PATENT ABSTRACT
An apparatus includes (a) switching power conversion circuitry including an inductive element connected to deliver energy via a unidirectional conducting device from an input source to a load during a succession of power conversion cycles, and circuit capacitance that can resonate with the inductive element during a portion of the power conversion cycles to cause a parasitic oscillation, and (b) clamp circuitry connected to trap energy in the inductive element and reduce the parasitic oscillation.

PATENT DESCRIPTION
BACKGROUND 
     This invention relates to reducing energy loss and noise in power converters. 
     As shown in  FIGS. 1 and 2 , in a typical PWM non-isolated DC-to-DC shunt boost converter  20  operated in a discontinuous mode, for example, power is processed in each of a succession of power conversion cycles  10 . During a power delivery period  12  of each power conversion cycle  10 , while a switch  22  is open, power received at an input voltage Vin from a unipolar input voltage source  26  is passed forward as a current that flows from an input inductor  21  through a diode  24  to a unipolar load (not shown) at a voltage Vout. Vout is higher than the input voltage, Vin. 
       FIGS. 2A and 2B  show waveforms for an ideal converter in which there are no parasitic capacitances or inductances and in which the diode  24  has zero reverse recovery time. During the power delivery period  12 , the current in the inductor falls linearly and reaches a value of zero at time tcross. At tcross, the ideal diode immediately switches off, preventing current from flowing back from the load towards the input source, and the current in the inductor remains at zero until the switch  22  is closed again at the next time ts 1 off. Thus, no energy is stored in the inductor  21  between times tcross and ts 1 on. 
     During another, shunt period  14  of each cycle, while switch  22  is closed, the voltage at the left side of the diode (node  23 ) is grounded, and no current flows in the diode. Instead, a shunt current (Is) is conducted from the source  26  into the inductor  21  via the closed switch  22 . In a circuit with ideal components, the current in the inductor would begin at zero and rise linearly to time ts 1 off, when switch  22  is turned off to start another power delivery period  12 . 
     In a non-ideal converter, in which there are parasitic circuit capacitances and the diode is non-ideal (e.g., for a bipolar diode there will be a reverse recovery period and for a Schottky diode there will be diode capacitance), an oscillatory ringing will occur after tcross. 
     In one example, waveforms for a non-ideal converter of the kind shown in  FIG. 1  are shown in  FIGS. 2C and 2D . Because of the reverse recovery characteristic of the diode, the diode does not block reverse current flow at time tcross. Instead, current flows in the reverse direction through the diode  24  and back into the inductor  21  during a period  18 . At time tdoff, the diode snaps fully off and the flow of reverse current in the diode goes to zero. 
     Because of the reverse flow of current in the diode during the diode recovery period, energy has been stored in the inductor as of the off time tdoff (the “recovery energy”). In addition, parasitic circuit capacitances (e.g., the parasitic capacitances of the switch  22 , the diode  24 , and the inductor  24 , not shown) also store energy as of time tdoff (e.g., the parasitic capacitance of switch  22  will be charged to a voltage approximately equal to Vout). 
     After time tdoff, energy is exchanged between the inductor and parasitic capacitances in the circuit. As shown in  FIGS. 2C and 2D , the energy exchange causes oscillatory ringing noise in the circuit. Furthermore, the presence of oscillatory current will generally result in energy being dissipated wastefully in the circuit at the start of the next shunt period when the switch is closed at time ts 1 on. The energy loss can amount to several percent of the total energy processed during a cycle. 
     SUMMARY 
     In general, in one aspect, the invention features apparatus that includes (a) switching power conversion circuitry including an inductive element connected to deliver energy via a unidirectional conducting device from an input source to a load during a succession of power conversion cycles, and circuit capacitance that can resonate with the inductive element during a portion of the power conversion cycles to cause a parasitic oscillation, and (b) clamp circuitry connected to trap energy in the inductive element and reduce the parasitic oscillation. 
     Implementations of the invention may include one or more of the following. The power conversion circuitry comprises a unipolar, non-isolated boost converter comprising a shunt switch. The power conversion circuitry is operated in a discontinuous mode. The clamp circuitry is configured to trap the energy in the inductor in a manner that is essentially non-dissipative. The clamp circuitry comprises elements configured to trap the energy by short-circuiting the inductor during a controlled time period. The inductive element comprises a choke or a transformer. The elements comprise a second switch connected effectively in parallel with the inductor. The second switch is connected directly in parallel with the inductor or is inductively coupled in parallel with the inductor. The second switch comprises a field effect transistor in series with a diode. 
     The power conversion circuitry comprises a unipolar, non-isolated boost converter comprising a shunt switch and a switch controller, the switch controller being configured to control the timing of a power delivery period during which the shunt switch is open and a shunt period during which the shunt switch is closed. 
     The shunt switch is controlled to cause the power conversion to occur in a discontinuous mode. The second switch is opened for a period before the shunt switch is closed in order to discharge parasitic capacitances in the apparatus. The power conversion circuitry comprises at least one of a unipolar, isolated, single-ended forward converter, a buck converter, a flyback converter, a zero-current switching converter, a PWM converter, a bipolar, non-isolated, boost converter, a bipolar, non-isolated boost converter, a bipolar, non-isolated buck converter, a bipolar, isolated boost converter, or a bipolar, isolated buck converter. 
     In general, in another aspect, the invention features, a method that reduces parasitic oscillations by trapping energy in the inductive element during a portion of the power conversion cycles. 
     Implementations of the invention include releasing the energy from the inductor essentially non-dissipatively. The energy is trapped by short-circuiting the inductive element during a controlled time period. The short-circuiting is done by a second switch connected effectively in parallel with the inductive element. The second switch is opened for a portion of the power conversion cycle in order to discharge parasitic capacitances. The invention reduces undesirable ringing noise generated in a power converted by oscillatory transfer of energy between inductive and capacitive elements in the converter and recycles this energy to reduce or eliminate the dissipative loss of energy associated with turn-on of a switching element in the converter. 
    
    
     
       Other advantages and features will become apparent from the following description and from the claims. 
       DESCRIPTION 
         FIG. 1  shows a power conversion circuit. 
         FIGS. 2A-2D  shows timing diagrams. 
         FIGS. 3 ,  5  and  6  show power conversion circuits with recovery switches. 
         FIG. 4  shows a timing diagram. 
         FIG. 7  shows a PWM, unipolar, isolated buck converter comprising a clamp circuit. 
         FIGS. 8A and 8B  show waveforms for the converter of FIG.  7 . 
         FIGS. 9A ,  9 B,  9 C, and  9 D show isolated, single-ended converters which comprise a clamp circuit. 
     
    
    
     With reference to  FIGS. 1 ,  2 C and  2 D, at time tdoff the parasitic capacitance across the switch  22  is charged to a voltage (approximately equal to Vout) which is greater than Vin and a current flows in L 1  owing to the reverse recovery of the diode  24 . 
     After tdoff, with the switch  22  open and the diode non-conductive, energy stored in the resonant circuit formed by the circuit parasitic capacitances and inductor L 1  causes oscillatory ringing in Iin and Vs. This oscillation (referred to herein as “parasitic oscillation” or simply “noise”) is unrelated to the power conversion process, and may require that noise filtering components be added to the converter (not shown). In addition, closure of the switch  22  after tdoff will result in a wasteful loss of some or all of this energy (“switching loss”). 
     By providing mechanisms for clamping the circuit voltages, the noise can be reduced or eliminated, and the stored energy can be trapped in an inductor and then released essentially losslessly back to the circuit. Generally, the capturing and later release of the energy is achieved by effectively shorting and then un-shorting the two ends of an inductor at controlled times. 
     As shown in  FIG. 3 , in one implementation, a unipolar, non-isolated, discontinuous boost converter circuit  28  includes a series circuit, comprising a recovery switch Rs  30  and a diode  32 , that is connected across the ends of the inductor  34 , and a controller  36  that regulates the on and off periods of both the recovery switch  30  and the shunt switch  22 . 
     The recovery switch  30  is turned on and off in the following cycle. The switch may be turned on any time during the power delivery period  12  when the voltage across the inductor, VB (FIG.  3 ), is negative, because this will result in diode  32  being reverse biased. During the reverse recovery period, the diode  32  prevents the current that is flowing backward from the diode  38  from flowing in recovery switch  30 . Instead, the reverse recovery energy is stored in the inductor. 
     After the diode snaps off, the energy stored in circuit parasitic capacitances will be exchanged with the inductor and the voltage, Vs, across shunt switch  22  will ring down. When the input voltage Vs rings down to the input voltage, Vin, the voltage VB will equal zero, the recovery diode  32  will conduct and the recovery switch  30  and the diode  32  will short the ends of the inductor  34 . In that state, the inductor  34  cannot exchange energy with any other circuit components. Therefore, the energy is “trapped” in the inductor and ringing in the main circuit is essentially eliminated. 
     Later, prior to the shunt switch being closed to start the shunt period, the recovery switch is opened. Because the current trapped in the inductor flows in the direction back toward the input source, opening the recovery switch  30  will result in an essentially lossless charging and discharging of parasitic circuit capacitances and a reduction in the voltage, Vs, across the shunt switch. By providing for a reduction in shunt switch voltage, Vs, the loss in the shunt switch associated with discharging of parasitics (“turn-on loss”) can be reduced or, in certain cases, essentially eliminated. 
     As shown in  FIG. 4 , the delay between the opening of the recovery switch  30  and the closing of the shunt switch  22  may be adjusted so that the closure of the shunt switch corresponds in time to approximately the time of occurrence of the first minimum in the voltage Vs following the opening of the recovery switch at time trsoff (the dashed line in the Figure shows how the voltage Vs would continue to oscillate after ts 1 on if the shunt switch  22  were not turned on at that time). In case where the voltage rings all the way down to zero (not shown in the Figure) the turn-on loss in the shunt switch can be essentially eliminated. Since capacitance energy is proportional to the square of the voltage, however, any amount of voltage reduction is important. 
     As shown in  FIG. 5 , in another approach, instead of wiring the recovery switch and diode directly across the inductor, a recovery switch  50  and a diode  52  are connected in series with a secondary winding  54  that is transformer-coupled to the inductor. The series circuit is connected to the ground side of the circuit for convenience in controlling the switch. The control switch may be implemented as a MOSFET in series with a diode. Turn-on losses will occur as a result of the body capacitor of the switch  50 , but they are relatively small because the switch die is relatively small. 
     As shown in  FIG. 6 , in another implementation, a bipolar discontinuous boost converter  60  operating from a bipolar input source, Vac, uses the transformer-coupled switching technique of  FIG. 5 , but includes two recovery switches  62 ,  64  connected to respective ends of the winding  66 . One of the recovery switches is always on for one polarity of input source Vac, and the other recovery switch is turned on and off using the same strategy as in FIG.  5 . The scenario is reversed when the polarity of the input source reverses. 
     Care must be taken not to have the shunt switch and the recovery switch on at the same time, which would short-circuit the source. 
     The energy-trapping technique may be applied to any power converter, isolated or non-isolated, PWM or resonant, in which energy storage in inductive and capacitive circuit elements results in parasitic oscillations within the converter. 
       FIG. 7 , for example, shows a PWM, unipolar, isolated buck converter  70  comprising a clamp circuit  76 . In such a converter, the voltage delivered by the input source  72 , Vin, is higher than the DC output voltage, Vout, delivered to the load  81 . In a first part of a converter operating cycle, the switch  74  is closed and energy is delivered to the load from the input source  72  via the output inductor  82 . In a second part of a converter operating cycle, the switch is open and energy stored in the inductor  82  flows as output current, Io, to the load via the diode  75 . For load values above some lower limit, the output current, Io, flows continuously in the output inductor Lout  82 . Below that lower limit, however, the instantaneous current in the output inductor  82  drops to zero and attempts to reverse. Under these circumstances the diode will block and, in the absence of the clamp circuit  76 , an oscillation will begin as energy is transferred back and forth between the inductor  82  and circuit parasitic capacitances (e.g., the parasitic capacitances of the switch  74 , the diode  75 , the inductor  82  and the clamp circuit  76 , not shown). Waveforms for the converter of  FIG. 7 , with the clamp circuit, are shown in  FIGS. 8A and 8B . 
     In  FIGS. 8A and 8B , the switch  74  is on at time t=0, the voltage VD is approximately equal to Vin, and the current Io is increasing owing to the polarity of the voltage impressed across Lout. At time tsoff, switch  74  turns off and the voltage VD drops to essentially zero volts as the parasitic capacitances across the diode  75  are discharged and the diode conducts. The clamp switch  78  may be turned on any time after the voltage VD drops below Vout. 
     At time tcross the current Io declines to zero and attempts to reverse. After the diode  75  ceases conducting, the voltage VD rings up until the clamp diode  80  begins to conduct at time tc, when the voltage VD is approximately equal to Vout. Between times tc and tcoff the clamp circuit clamps the inductor and prevents parasitic oscillations. At time tcoff, the clamp switch is opened and the voltage VD rings up toward Vin. At time tson the switch  74  is closed, initiating another converter operating cycle. A switch controller  77  controls the relative timing of the two switches  74 ,  78 . As for the timing discussed in  FIG. 4 , the delay between the opening of the clamp switch  78  and the closing of the switch  74  is adjusted so that the closure of switch  74  corresponds in time to approximately the time of occurrence of the first maximum in the voltage Vs following the opening of the clamp switch  78 . This minimizes or eliminates the switching loss associated with closure of switch  74 . 
     The transformer coupled clamp circuit of  FIG. 5  may be used in the converter of FIG.  7 . 
     Other embodiments are within the scope of the following claims. 
     For example, the technique may be applied to any switching power converter in which there is a time period during which undesired oscillations occur as a result of energy being transferred back and forth between unclamped inductive and capacitive energy storing elements. 
     For example,  FIGS. 9A through 9D  show isolated, single-ended converters which comprise a clamp circuit  76  according to the invention.  FIG. 9A  is a unipolar, single-ended, forward PWM converter;  FIG. 9B  is a unipolar, single-ended, zero-current switching forward converter (as described in U.S. Pat. No. 4,415,959, incorporated by reference);  FIG. 9C  is a unipolar, single-ended, flyback converter with a clamp circuit  76  connected to the primary winding  105  of the flyback transformer; and  FIG. 9D  is a unipolar, single-ended, flyback converter with a clamp circuit  76  connected to the secondary winding  104  of the flyback transformer. 
     The clamp circuit may be modified to be of the magnetically coupled kind shown in  FIG. 5 , above. Other topologies to which the technique may be applied include resonant and quasi-resonant non-isolated, boost, buck and buck-boost converters. By use of bipolar clamp circuitry of  FIG. 6 , or equivalent circuitry, the technique may be applied to bipolar equivalents of unipolar PWM, resonant and quasi-resonant non-isolated, boost, buck and buck-boost converters.