PATENT ABSTRACT
An apparatus comprising a functional circuitry on a first die. Said function circuitry configured to drive an RF voltage isolation link with an RF signal responsive to receipt of a logic signal at a first logic state. Control circuitry modifies the frequency of the RF signal to spread harmonics to other than a fundamental frequency.

PATENT DESCRIPTION
CROSS-REFERENCE TO RELATED APPLICATIONS 
       [0001]    This application is a continuation of U.S. patent application Ser. No. 11/945,914, filed on Nov. 27, 2007, entitled SPREAD SPECTRUM ISOLATOR, issued Jan. 19, 2010 as U.S. Pat. No. 7,650,130, which is a continuation of U.S. Pat. No. 7,302,247, entitled SPREAD SPECTRUM ISOLATOR, issued Nov. 27, 2007; which is a continuation-in-part of U.S. Pat. No. 7,421,028, entitled TRANSFORMER ISOLATOR FOR DIGITAL POWER SUPPLY, issued Sep. 2, 2009, U.S. Pat. No. 7,447,492, entitled ON-CHIP TRANSFORMER ISOLATOR, issued Nov. 4, 2008, U.S. Pat. No. 7,376,212, entitled RF ISOLATOR WITH DIFFERENTIAL INPUT/OUTPUT, issued May 20, 2008 and U.S. Pat. No. 7,460,604, entitled RF ISOLATOR FOR ISOLATING VOLTAGE SENSING AND GATE DRIVERS, issued Dec. 2, 2008. 
     
    
     TECHNICAL FIELD 
       [0002]    The present invention relates to digital isolators, and more particularly, to digital isolators providing isolation for voltage sensing and gate drivers. 
       BACKGROUND 
       [0003]    Within power conversion products, there is a need for high speed digital links that provide high isolation at a low cost. Typical digital links within power conversion products require a speed of 50-100 megabits per second. Isolation between the input and output of power conversion products is required in the range of 2,500-5,000 V. Existing solutions for providing a high speed digital isolation link have focused on the use of magnetic pulse couplers, magnetic resistive couplers, capacitive couplers and optical couplers. 
         [0004]    Referring now to  FIG. 1 , there is illustrated the general block diagram of a system using a magnetic pulse coupler to isolate a digital link  102  between a driver  104  and a detector  106 . The driver  104  resides upon one side of the digital link  102  and transmits information over the digital link  102  to the detector  106  residing on the other side of the digital link. Resting between the driver  104  and detector  106  is a pulse transformer  108 . The pulse transformer  108  provides an electromagnetically coupled transformer between the driver  104  and detector  106 . The pulse transformer  108  generates a pulse output in response to a provided input from the driver as illustrated in  FIG. 2 . The input from the driver  104  consists of the two pulses  202  and  204 . Each pulse  202 ,  204  consists of a rising edge  206  and a falling edge  208 . In response to a rising edge  206 , the output of the pulse transformer  108  generates a positive pulse  210 . The falling edge  208  of a pulse generates a negative pulse  212 . The pulse transformer circuit illustrated with respect to  FIGS. 1 and 2  suffers from a number of deficiencies. These include start-up where the detector  106  will not know at what point the input from the driver has begun, whether high or low until a first edge is detected. Additionally, should any error occur in the pulse output of the pulse transformer  108 , the detector  106  would have a difficult time determining when to return to a proper state since there may be a long period of time between pulses. 
         [0005]    Referring now to  FIG. 2 , there is illustrated an alternative prior art solution making use of a magneto resistive coupler. The magneto resistive coupler  302  consists of a resistor  304  and associated transformer  306 . The resistor  304  has a resistance value that changes responsive to the magnetic flux about the resistor. The transformer detector  306  utilizes a wheatstone bridge to detect the magnetic flux of the resistor and determined transmitted data. 
         [0006]    Another method of isolation between a driver  404  and a detector  406  is illustrated in  FIG. 4 . The driver  404  and the detector  406  are isolated on opposite sides of a digital link  402  by a capacitor  408 . The capacitor  408  capacitively couples the driver  404  and detector  406  together to achieve a level of isolation. A problem with the use of capacitive coupling to isolate digital links is that capacitive coupling provides no common mode rejection. 
         [0007]    An additional problem with some isolator designs involves the reception of RF interference from nearby transmitting GSM, DCS and CDMA cellular telephones. The problem is caused by the application printed circuit board acting as a dipole antennae at GHz frequencies. This results in large common mode signals being seen at the isolator at RF frequencies. Some manner for minimizing these large common mode signals at GHz frequencies would be highly desirable. 
         [0008]    Thus, an improved method for providing isolation over high speed digital links within power supply components would be greatly desirable. 
       SUMMARY 
       [0009]    The present invention disclosed and claimed herein, in one aspect thereof, comprises functional circuitry on a first die. Said function circuitry configured to drive an RF voltage isolation link with an RF signal responsive to receipt of a logic signal at a first logic state. Control circuitry modifies the frequency of the RF signal to spread harmonics to other than a fundamental frequency. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0010]    For a more complete understanding, reference is now made to the following description taken in conjunction with the accompanying Drawings in which: 
           [0011]      FIG. 1  illustrates a block diagram of a prior art magnetic pulse coupler isolator; 
           [0012]      FIG. 2  illustrates the input and output signals of the prior art magnetic pulse transformer of  FIG. 1 ; 
           [0013]      FIG. 3  illustrates a prior art magneto resistive coupler; 
           [0014]      FIG. 4  illustrates a prior art capacitive coupler; 
           [0015]      FIG. 5  illustrates a switched power supply including isolation circuitry; 
           [0016]      FIG. 6  illustrates an RF isolation link of the present disclosure; 
           [0017]      FIG. 6   a  illustrates a schematic block diagram of a circuit for providing the RF isolation link using frequency modulation; 
           [0018]      FIG. 7  illustrates a schematic diagram of the circuitry for providing the RF isolation link using amplitude modulation; 
           [0019]      FIG. 8  illustrates the waveforms present on the transmit side of the RF isolation link of  FIG. 7 ; 
           [0020]      FIG. 9  illustrates the waveforms present on the receiving side of the RF isolation link of  FIG. 7 ; 
           [0021]      FIG. 10  illustrates the frequency response of the RF isolation link; 
           [0022]      FIG. 11  illustrates a model of one of the transformers included within the RF isolation link; 
           [0023]      FIG. 12  illustrates the frequency response of one transformer of the RF isolation link; 
           [0024]      FIG. 13  illustrates the voltages across each transformer included within an RF isolation link and across the entire RF isolation link; 
           [0025]      FIG. 14   a  is a block diagram illustrating the circuitry included within a chip on one side of an RF isolation link for providing multiple isolation link channels; 
           [0026]      FIG. 14   b  is a schematic diagram of an oscillator circuit; 
           [0027]      FIG. 14   c  is a block diagram of the logic circuit of  FIG. 17   a;    
           [0028]      FIG. 15  illustrates a pair of chips within a single package including four separate channels for providing four isolated digital data links; 
           [0029]      FIG. 15   a  illustrates the RF isolation link within a chip package; 
           [0030]      FIG. 16  illustrates an integrated RF isolation link in a single package including two dies; 
           [0031]      FIG. 16   a  illustrates an integrated RF isolation link in a single package having a digital input and a digital output; 
           [0032]      FIG. 16   b  illustrates an integrated RF isolation link in a single package including a digital input/output and an analog input/output; 
           [0033]      FIG. 16   c  illustrates an integrated RF isolation link in a single package including an analog input/output and an analog input/output; 
           [0034]      FIG. 17   a  illustrates an RF isolation link integrated with a microcontroller; 
           [0035]      FIG. 17   b  illustrates an RF isolation link integrated with a microcontroller interconnected to a second chip providing both analog input and analog output; 
           [0036]      FIG. 18   a  illustrates one coil of a transformer of the RF isolation link; 
           [0037]      FIG. 18   b  illustrates a second coil of a transformer of the RF isolation link; 
           [0038]      FIG. 19  illustrates an overlapping view of the transformers described in  FIGS. 21   a  and  21   b;    
           [0039]      FIG. 20  is a side view of the coils forming a transformer of the RF isolation link; 
           [0040]      FIG. 21  illustrates an offset between metal layers to increase breakdown voltages within a transformer; 
           [0041]      FIG. 22  illustrates a perspective cutaway view of the coil construction; 
           [0042]      FIG. 23  illustrates the separate areas in which the transformer coils and the circuitry would be implemented on a chip utilizing the RF isolation circuit; 
           [0043]      FIG. 24  illustrates the structure of the RF isolation link integrated on a single chip; 
           [0044]      FIG. 25  illustrates an isolator chip having two printed circuit boards which may act as a dipole antenna at higher frequencies; 
           [0045]      FIG. 26  illustrates the parasitic capacitance between windings at higher frequencies; 
           [0046]      FIG. 27  illustrates how RF signals can be passed through the transformer as common mode signals; 
           [0047]      FIG. 28  illustrates a schematic diagram of an RF isolator including a differential output; 
           [0048]      FIG. 29  is a more detailed schematic view of the RF isolator of  FIG. 28 ; 
           [0049]      FIG. 30  is an illustration of the transformer coils of the RF isolator including a center tap; 
           [0050]      FIG. 31  illustrates the manner in which voltage may be altered to maintain optimized receiver/transmitter gain; 
           [0051]      FIG. 32  is a schematic diagram illustrating a prior art method for generating the band gap reference voltage; 
           [0052]      FIG. 33  is a schematic diagram of the manner for generating the band gap reference voltage according to the present disclosure; 
           [0053]      FIG. 34  illustrates a prior art method for generating a reference voltage; 
           [0054]      FIG. 35  illustrates a modified method for generating a band gap reference voltage; 
           [0055]      FIG. 36  illustrates a switched power supply including a PWM controller and power transistors; 
           [0056]      FIG. 37  illustrates a prior art method for isolating a PWM controller on a primary side from drivers on a secondary side of a switched power supply; 
           [0057]      FIG. 38  illustrates a second prior art method for isolating a PWM controller on a primary side from a driver circuit on a secondary side of a switched power supply; 
           [0058]      FIG. 39  illustrates a final prior art embodiment for isolating a PWM controller on a primary side from a driver on a secondary side of a switched power supply; 
           [0059]      FIG. 40  illustrates a block diagram of an isolated gate driver for voltage isolating a PWM controller from power transistor circuitry; 
           [0060]      FIG. 41  is a general schematic diagram of the isolated gate driver; 
           [0061]      FIG. 42  illustrates a circuit package including two separate dies for implementing the isolated gate driver; 
           [0062]      FIG. 43  is a detailed schematic diagram of the circuitry for implementing the isolated gate driver; 
           [0063]      FIG. 44  is a schematic diagram of the level shifter of  FIG. 41 ; 
           [0064]      FIG. 45  illustrates a prior art method for isolating a voltage sensing circuit from a PWM controller; 
           [0065]      FIG. 46  is a schematic block diagram of a method for isolating voltage sensing between an output voltage on a secondary side and a PWM controller on a primary side; 
           [0066]      FIG. 47  illustrates an integrated chip including circuitry for voltage isolating gate drivers from PWM controllers on primary and secondary sides of a switched power supply and for voltage isolating the voltage sensing function on a secondary side from a PWM controller on a primary side of a switched power supply; 
           [0067]      FIG. 48   a  illustrates the use of a single RF frequency for use with the RF isolator; 
           [0068]      FIG. 48   b  illustrates the radiated emissions for an RF isolator using a single RF frequency; 
           [0069]      FIG. 49   a  illustrates the use of a stepped frequency that steps between a first and second frequency; 
           [0070]      FIG. 49   b  illustrates the radiated emissions of the stepped frequency RF isolator; 
           [0071]      FIG. 50  illustrates a block diagram of a first embodiment of a circuit for generating a stepped RF carrier signal; 
           [0072]      FIG. 51  is a schematic diagram of the RF oscillator circuit used in the circuit of  FIG. 50 ; 
           [0073]      FIG. 52  illustrates a schematic diagram of the slow oscillator circuit of  FIG. 50 ; 
           [0074]      FIG. 53  illustrates a block diagram of a second embodiment of a circuit for generating a stepped RF carrier signal; 
           [0075]      FIG. 54  is a schematic diagram of the circuit of  FIG. 52 ; 
           [0076]      FIG. 55  illustrates the modeled results for the circuit of  FIG. 50 ; and 
           [0077]      FIG. 56  illustrates an alternative embodiment for creating a random code which is used for generating the stepped RF carrier signal. 
       
    
    
     DETAILED DESCRIPTION 
       [0078]    Referring now to the drawings, and more particularly to  FIG. 5 , there is illustrated a block diagram of a DC-DC switching power supply utilizing an RF isolation link. Switching power supplies utilize a plurality of switches which are turned on and off to switch an input DC voltage across a transformer to a load, the output voltage at a different DC voltage level. By switching the current inductively coupled through the transformer to the load in a particular manner, a DC output voltage at a different voltage level than the input DC voltage can be provided to the load. The controlled switching is typically facilitated with some type of control circuit. This control circuit can be an analog control circuit formed from a plurality of analog discrete devices, or it can be a digital circuit. In digital control circuits, digital signal processors (DSPs) and microcontroller units (MCU) have been utilized. The DSPs control the duty cycle and relative timing of the switches such that the edges of each control pulse to the various transistor switches controlling power delivery to the load is varied. In order to perform this operation in the digital domain, the DSP must perform a large number of calculations, which requires a fairly significant amount of code to be generated to support a specific power supply topology, operating frequency, component characteristics and performance requirements. For example, inductor size decreases with increasing PWM frequency, dead times increase with increasing transistor turn-off times, and so on. Although DSPs can handle the regulation tasks, they are fairly complex and expensive and code changes in power supply applications are difficult. 
         [0079]    Referring further to  FIG. 5 , the power supply includes a primary switch group  502  that is operable to receive an input voltage on a node  504 , this being a DC voltage, and ground on a node  506 . The primary switch group  502  is coupled through an isolation transformer  508  to a secondary switch group  510 . The secondary switch group  510  is operable to drive an input voltage node  512  that is connected to one terminal of a load  514 , the secondary switch group  510  also having a ground connection on a node  516 , the load  514  disposed between the node  512  and the node  516 . The two switch groups  502  and  510  are operable to operate in conjunction with various pulse inputs on a control bus  518  associated with the primary switch group  502  and with various pulse inputs on a control bus  526  associated with the secondary switch group  510 . 
         [0080]    A digital control circuit  524  is provided for controlling the operation of the primary switch group  502  and the secondary switch group  510 . The voltages on nodes  504  and  506  are provided as inputs to the digital control circuit  524  for sensing the voltage and current on the primary side, the digital control circuit  524  generating the information on the bus  518  for control of the primary switch group  502 . The control circuit  524  must be isolated from the secondary group switch  510 , since there can be a significant DC voltage difference therebetween. This is facilitated by driving the bus  526  through an isolation circuit  528 , such as the RF isolation circuit which will be discussed herein below, to drive the bus  520 . Similarly, the control circuit  524  is operable to sense the voltage and current levels on the output node  512  through sense lines  530  which are also connected through an isolation circuit  532  to the digital control circuit  524 . The digital control circuit  524  is also interfaced to a bus  536  to receive external control/configuration information. This can be facilitated with a serial databus such as an SMB serial databus. 
         [0081]    Referring now to  FIG. 6 , there is illustrated the RF isolation link of the present disclosure. The RF isolation link  600  of the present disclosure is implemented by integrating a portion of the link in two chips or dies between which a high rate data link with voltage isolation is required. Each chip  602  includes a transformer  604  and transmit and receive circuitry  606  for providing the RF isolation link  600  between the chips. Alternatively, the chip  602  could include only transmit circuitry or receive circuitry with the partnered chip, including a corresponding receiver or transmitter. The RF signals are generated within the transmit/receive circuitry  606  on one side of the RF isolation link, and the RF signals are transmitted between the chips  602  utilizing the transformers  604  in each chip and the magnetic coupling effect therebetween. 
         [0082]    Once the RF signals are received at the receiving side, the transmit and receive circuitry  606  detects the data contained within the transmission from the first chip and utilizes the data as appropriate. While the description with respect to  FIG. 6  only illustrates the transformer  604  and transmit and receive circuitry  606  within each chip  602 , additional circuitry will be implemented on the chips  602  for performing processing functions associated with the data transmitted over the RF isolation link  600 . The data transmitted over the RF isolation link  600  may be transmitted using either frequency modulation techniques or amplitude modulation techniques. In the preferred embodiment of the disclosure, discussed with respect to  FIG. 7  herein below, AM modulation is used for transmitting the data. 
         [0083]    In operation, each of the transmit/receive circuits  606  operates in either transmit or receive mode. In the transmit mode, digital data received on a digital bus  603  is serially transmitted from one of the transmit/receive circuit  606  to the other one on the other of the dies  602 . This is facilitated by driving the transformer  606  with a signal such that energy is coupled from the primary to the secondary thereof. This will allow energy to be transmitted on transmission lines  605  that couple the transformers  604  together. Each of the transformers is comprised of a primary  607  and a secondary  609 . The primary  607  is driven with the input signal and energy associated therewith is coupled across the high voltage isolation boundary from the primary  607  to the secondary  609  and onto the transmission line  605 . As will be described herein below, both of the transmit/receive circuits  606  and transformers  604  are fabricated on an integrated circuit such that the primary  607  and secondary  609  are both formed thereon utilizing conventional processing techniques and available conductive layers that are shared with the transmit/receive circuits. There will be a loss associated with the coupling coefficient between the primary and secondary such that the amount of energy that can be delivered from the transmit/receive circuit  606  to the transmission line  605  is reduced and, further, there will be more loss at certain frequencies than others. As such, the transformer  604  will have a unique frequency response where the loss will be greater at some frequencies than others. To accommodate this, the transmit/receive circuit  606  has contained therein a transmitter operating at a defined frequency that is within the lowest loss portion of the frequency response of the transformer  604 . By utilizing various modulation schemes, data can be transmitted on this carrier to the transmission line  605 . The operation of the transmitter/receiver circuit  606  will be described in more detail herein below. 
         [0084]    Referring now to  FIG. 6   a , there is illustrated an alternate embodiment of the switching power supply utilizing frequency modulation to transmit data between a pair of chips over an RF isolation link  600 . The description with respect to  FIG. 6   a  is merely provided as an illustration of one potential embodiment of an FM circuit used for creating an RF isolation link, and one skilled in the art would realize the possibility of numerous additional embodiments. The data is input on a data bus  610  into a Manchester encoding circuit  612 , a conventional data encoding circuit. Also input to the Manchester encoding circuit  612  is a clock signal. The clock signal is also input to a voltage controlled oscillator  614 . Data is output from the Manchester encoding circuit  612  and applied to a divide circuit  616 . A second input of the divide circuit  616  is connected to the output of the voltage controlled oscillator  614 . The output of the divide circuit  616  is connected to a second input of the voltage controlled oscillator  614  to allow modulation thereof with the Manchester encoding circuit  616 . The voltage controlled oscillator  614  outputs a frequency modulated signal representing the received data on bus  610  to a driver  618 . The signal is filtered by a capacitor  620  before being applied to a transformer  622 . The FM modulated signal is coupled by transformer  622  onto transmission lines  624  passing across an interface  626  between either a first and second chip that are to be voltage isolated from each other. 
         [0085]    The received data signal is electromagnetically coupled onto the receiver circuitry by a second transformer  628 . The received signal passes through a limiter circuit  630  whose output is applied to a Divide-by-N circuit  632  and a discriminater circuit  634 . The output of the Divide-by-N circuit  632  is applied to the input of a PFD (phase/frequency detector) circuit  636 . A second input to the PFD circuit  636  is provided by a second Divide-by-N circuit  638  having its input connected to the output of the voltage controlled oscillator  640 . The input of the voltage controlled oscillator  640  is connected to the output of the PFD circuit  636 . The output of the voltage controlled oscillator  640  is connected to a second input of the discriminater  634 , this being a phase locked output phase locked to the data clock. The discriminater circuit  634  determines the data contained within the received signal responsive to the output of the voltage controlled oscillator  640  and the limiter  630 . This data is provided to a latch circuit  636  having its clock input connected to the output of the Divide-by-N circuit  638 . The data output of the receiver is provided from the latch circuit  642 . 
         [0086]    Referring now to  FIG. 7 , there is illustrated the preferred embodiment of the RF isolation link  600  of the present disclosure wherein amplitude modulation is used to transmit data over the link. The RF isolation link  600  consists of transmitter circuitry  702  and receiver circuitry  704 . The transmitter circuitry  702  consists of a NAND gate  708  having a first input connected to receive the data to be transmitted over the RF isolation link  600  and a second input connected to receive the RF carrier signal. The RF carrier in the preferred embodiment comprises a 2 GHz signal. The data input to the first input of the NAND gate  708  consists of either a logical A1@ or A0@ which will selectively gate the RF carrier signal to the output of NAND gate  708  in the presence of a logical A1.@ This causes the output  709  of the NAND gate  708  to either provide the RF carrier signal when the data bit is A1,@ or not provide the RF signal when the data bit is “0.” The output of the NAND gate  709  is connected to the gate of a p-channel transistor  710 . The drain-source path of the p-channel resistor  710  is connected between V DD  and ground through a resistor  712  and a first transformer  714 . The transformer  714  electromagnetically couples the RF carrier signal to transformer  718  via lines  716 . This links the data represented by the RF carrier signal between the first chip  602   a  and the second chip  602   b  while providing voltage isolation between the chips  602  via the first and second transformers  714 ,  718 . Each of the transformers  714  and  718  are associated with a particular chip  602  on opposite sides of interface  720 . Thus, wherein previous systems required a separate chip to provide an isolation link between two separate chips, the present disclosed device integrates the RF isolation link  600  onto the chips  602 . 
         [0087]    The receiver circuitry  704  receives the signal which has been electromagnetically coupled via transformer  714  onto the transmission lines  716  to transformer  718 . The receiver circuit  704  consists of an amplifier  705  and a detector  706 . The amplifier  705  provides two stages of amplification consisting of a first amplification stage including a capacitor  722  in series with an amplifier  724  and a feedback resistor  726 . The second amplifier stage is similar to the first amplifier stage and includes a capacitor  728  in series with an amplifier  730  and a feedback resistor  732 . These two stages amplify the received signal from the transformer  718 . 
         [0088]    The detector  706  detects the presence or absence of the RF carrier signal within the amplified received signal to determine the data being transmitted from the first chip  602   a . The amplified signal from the amplifier  705  is first filtered by a capacitor  734 . N-channel transistor  736  has the gate thereof connected to capacitor  734  and has the source-drain path thereof connected to one side of a current mirror comprised of p-channel transistors  738  and  740 . The source-drain path of transistor  738  is connected between V DD  and node  742 , the gate thereof connected to the gate of transistor  740 . The source-drain path of transistor  740  is connected between V DD  and a node  743 , the gate thereof connected to node  743  to provide a diode connected configuration. The output of the detector  706  is provided from node  742  at which the source-drain path of the n-channel transistor  736  is connected to the p-channel transistor  738  of the current mirror. A bias network is provided by n-channel transistors  744  and  746  which have the source-drain paths thereof connected between node  743  and ground and the gates thereof connected to a node  745  through a resistor  748 , with a capacitor  750  connected between node  745  and ground. Biasing is also provided by resistor  752  connected between node  745  and the gate of transistor  736 , a diode connected p-channel transistor  754  connected between node  745  and ground and a current source  756  for driving node  745 . When no RF signal is detected by the receiver, the Data Out from node  742  of the detector circuit  706  will be equal to V DD  since the PMOS current is greater than 1.33 times the NMOS current and a logical A0@ is detected. In the presence of the RF signal, the Data Out from node  742  will vary in response to the variation of the detected RF carrier signal and a logical A1.@ The detector  706  outputs a low voltage when RF is present and a high voltage when RF is absent relying on the nonlinear (square root) behavior of the MOS device directed by the alternating current. 
         [0089]    Referring now to  FIGS. 8 and 9 , there are illustrated the waveforms and data provided at the transmission side ( FIG. 8 ) of an RF isolation link  600  and the receive side ( FIG. 9 ) of the RF isolation link. On the transmission side illustrated in  FIG. 8 , the data  800  is either transmitted as a one bit (high) or zero bit (low). A one bit pulse is indicated at  802 ,  804  and  806 . A zero bit pulse is indicated at  808  and  810 . The transmit data provided to the transformer  714  is illustrated by the waveform  812 . The transmit data waveform represents the 2 GHz RF carrier signal. When a logical A1@ data bit is being transmitted and the data signal is high, the presence of the 2 GHz RF carrier is provided at the transmit data output. When a logical A0@ bit is being transmitted, the signal is virtually zero at the transmit data output. Thus, whether a logical A1@ bit or a logical A0@ bit is transmitted is indicated either by the presence or absence of the 2 GHz RF carrier signal. 
         [0090]      FIG. 9  illustrates the waveforms associated with the receiver  704 . The received data for the logic A1@ bit is represented at points  902 ,  904  and  906  and indicates the three 2.5 GHz 
         [0091]    RF carrier pulses transmitted from the transmitter  702  of the RF isolation link  600 . The received pulses are amplified by the amplifier  705  such that when the signal is input to the detector circuit  706 , the pulses are represented by the amplified waveform pulses  908 ,  910  and  912 . As discussed previously, the detector data output rises to V DD  at points  916 ,  918  when no RF carrier signal is detected by the detector  706  indicating a logical A0.@ When an RF carrier signal is detected, the output of the detector  706  begins to vary and drops low at points  920 ,  922  and  924  indicating a logical A1,@ this being the result of an increase in the NMOS current in transistor  736 . 
         [0092]    Referring now to  FIG. 10 , there is illustrated the frequency response of a channel having the RF isolation circuit  600  described in  FIG. 7 . 
         [0093]    Referring now to  FIG. 11 , there is illustrated a model for the transformers ( 714 ,  718 ) illustrated in  FIG. 7 . The input of the transformer consists of nodes  1002  and  1100 . Node  1002  is connected to ground through capacitor  1104  and resistor  1106 . Node  1100  is connected to ground through capacitor  1116  and resistor  1118 . Node  1102  interconnects with node  1100  via a parallel connection of capacitor  1108  in series with resistor  1110  and inductor  1112  in series with resistor  1114 . The output of the transformer consists of nodes  1122  and  1124 . Node  1122  is connected to ground through capacitor  1126  and resistor  1128 . Node  1124  is connected to ground through capacitor  1130  and resistor  1132 . Node  1122  interconnects with node  1124  via a parallel connection of capacitor  1134  in series with resistor  1136  and inductor  1138  in series with resistor  1140 . Nodes  1102  and  1122  are interconnected via a capacitor  1142  with a value of approximately 125 Ff. Nodes  1100  and  1124  are interconnected via a capacitor  1144  with a value of approximately 125 Ff. 
         [0094]    With specific reference to  FIG. 13 , it can seem that the low frequency response of the transformers is relatively lossy whereas the peak of the response occurs around 2.5 GHz. This is due to the manner in which the transformer was fabricated. Each side of the transformer is comprised of an inductive element, each inductive element on either side of the transformer coupled together through a layer of dielectric material, as will be described herein below. The series inductance value will result in an effect on the frequency response that will somewhat narrow the frequency response thereof. The amount of energy that is coupled from the output is a function of the coupling coefficient. The two sides of the transformers are disposed on a substrate, as will be described herein below, such that one element is disposed over the other element and separated therefrom by a high voltage dielectric to increase the effective breakdown voltage. This will allow high frequency energy to be coupled from one conductive element to the other. The voltage breakdown is a function of the properties of the material disposed between the two conductors at DC and the distance by which the two are separated. If the transformer were fabricated on a single layer of material in the semiconductor substrate, then the distances between the edges thereof would define the voltage breakdown. For example, the transformer device could be fabricated with the use of a directional coupler, which would provide a more broadband response. However, the area for such a design could be significant. 
         [0095]    It can be seen that, due to the low frequency attenuation of the transformer, it would be difficult to couple through energy from a DC pulse, since only the high frequency energy would be passed there through. As such, the spectral energy that is coupled through the transformer of the present disclosure is concentrated therein with the use of a high frequency carrier that is disposed substantially within the center of the frequency response of the transformer. This will allow a large portion of the energy generated to be coupled across the transformer. 
         [0096]    Using the RF isolation links  600  described above, voltage isolation of up to 5,000 volts may be achieved, 2,500 volts for each side. Thus, as illustrated in  FIG. 16 , the RF isolation circuit  602  may provide 5,000 volts of isolation between a first chip  602   a  and a second chip  602   b . While the voltage between the input terminals of the chip  602   a  will be zero volts, and the voltage between the input terminals of the chip  602   b  will also be zero volts, the total voltage difference between the two chips may be 5,000 volts with a 2,500 voltage difference across each of the transformers  714 ,  718  associated with the interfaces to the RF isolation circuit on each chip  602 . 
         [0097]    Referring now to  FIG. 14   a , there is illustrated a block diagram of the structure of an interface of a single chip  602  including a portion of a plurality of channels  1402  including the RF isolation link of the present disclosure. Each channel  1402  consists of the transformer  1406  and transmit and/or receive circuitry described with respect to  FIG. 7 . Data may be either input or received at the interface  1404  of transformer  1406 . Each channel  1402  is interconnected with a pad driver  1408  that either drives transmitted data from the pad driver over channel  1402  to be output over the interface  1404  or drives received data to the associated pad of the chip  602 . The manner in which data can be either transmitted or received over a particular channel  1402   a  is controlled on the chip  602  by logic circuitry  1410  providing control over various control lines  1412 . The manner in which the logic control  1410  controls whether a channel is used for transmitting or receiving is set by input bond pad options  1414 . Thus, in this embodiment, data is received as either a logic A1@ or a logic A0@ and the associated transformer is driven, when a pad is configured as a transmitter, (or not driven) accordingly. For received data on the associated transformer, when configured to receive data, the output of the pad is either high or low. 
         [0098]    An oscillator circuit  1430  is also associated with all of the channels of the interface. A band gap generator  1420  is provided on-chip and connected to V DD  to provide a band gap reference voltage to a regulator circuit  1422 . While the description with respect to  FIG. 14   a  only illustrates a single voltage regulator  1422 , it will be noted that a separate voltage regulator  1422  will be associated with each of the channels of the interface for noise purposes. The voltage regulator  1422  consists of an amplifier  1424  having one input connected to the output of the band gap generator  1420 . The output of the amplifier  1424  is connected to the gate of a transistor  1426 . The drain-source path of the transistor  1426  is connected between V DD  and a node  1427 . Node  1427  is also connected to the second input of the differential amplifier  1424 . A capacitor  1428  is connected between node  1422  and ground. Each of the channels  1402   a ,  1402   b ,  1402   c  and  1402   d  has a regulator  1422  associated therewith. Connected to node  1427  is an oscillator circuit  1430 . 
         [0099]      FIG. 14   b  illustrates the oscillator circuit  1430  of  FIG. 14   a . The output  1435  is connected to node  1437  between transistor  1436  and transistor  1438 . The drain-source path of transistor  1436  is connected between V DD  and node  1437 . The drain-source path of transistor  1438  is connected between node  1437  and ground. The gates of transistor  1436  and  1438  are connected to each other through a node  1439 . A transistor  1440  has its gate connected to ground and its drain-source path connected between V DD  and the gate of transistor  1440 . Node  1439  also interconnects transistor  1442  and transistor  1444 . The drain-source path of transistor  1442  is connected between V DD  and node  1439 . The drain-source path of transistor  1444  is connected between node  1439  and ground. The gates of transistors  1442  and  1444  are interconnected with each other via node  1445 . A capacitor  1446  is connected between node  1445  and ground. Node  1445  is connected to a first terminal of coil  1450 . The second terminal of coil  1450  interconnects with the circuit via node  1460 . Transistors  1452  and  1454  are interconnected via node  1445 . The drain-source path of transistor  1452  is connected between V DD  and node  1445 . The drain-source path of transistor  1454  is connected between node  1445  and ground. The gates of both transistor  1452  and  1454  connect to node  1460 . Transistors  1458  and  1456  are interconnected via node  1460 . The drain-source path of transistor  1458  is connected between V DD  and node  1460 . The drain-source path of transistor  1456  is connected between node  1460  and ground. The gates of transistors  1458  and  1456  connect to node  1445 . The capacitor  1462  is connected between node  1460  and ground. Also connected to node  1460  are the gates of transistors  1464  and  1466 . The drain-source pathway of transistor  1464  is connected between V DD  and node  1465 , and the drain-source pathway of transistor  1466  is connected between node  1465  and ground. This oscillator therefore comprises a conventional LC oscillator. 
         [0100]    Referring now to  FIG. 14   c , there is illustrated one embodiment of the circuitry which might be incorporated within the logic circuit  1410 . In this embodiment, the logic circuit  1410  includes of a decoder  1432 . The decoder has a total of three bond pad inputs B 0 , B 1  and B 2  for receiving the indication of the version of the chip being implemented. The outputs  1434  of the decoder are input to the appropriate channels such that the channel may be configured in either a transmission or reception mode. 
         [0101]    Referring now also to  FIG. 15 , there is illustrated the manner in which the single chip design described in  FIG. 16  can be used to facilitate an entire RF isolation circuit including four separate RF isolated channels. A first chip  1502  is reversed such that the output channels  1402  between the first chip  1502  and the second chip  1504  are merely reversed. Thus, when viewing the chip  1502  from top to bottom of chip one, channel one is at the top, channel two is second, channel three is third and channel four is last. For the second chip  1504 , the channels run in the opposite direction with channel one beginning at the bottom and channel four being at the top. The physical design of chip  1502  and chip  1504  are the same. Chip  1504  is merely reversed to facilitate the three versions of the chip as described below. Three different bond option versions may be selected for input to the logic circuit  1410  of the package containing the first chip  1502  and the second chip  1504  utilizing the decoder circuit  1432 . Referring now to the Table 1, there are illustrated the three separate versions of operation for both the first chip  1502  and the second chip  1504  and the indication of whether the channel comprises a transmit or receive channel in the associated version. 
         [0000]    
       
         
               
               
               
               
               
               
               
             
           
               
                   
                 TABLE 1 
               
               
                   
                   
               
               
                   
                 Version 
                 Chip 
                 Ch. 1 
                 Ch. 2 
                 Ch. 3 
                 Ch. 4 
               
               
                   
                   
               
             
             
               
                   
                 1 
                 1 
                 Tx 
                 Tx 
                 Tx 
                 Tx 
               
               
                   
                 2 
                 1 
                 Tx 
                 Tx 
                 Rx 
                 Rx 
               
               
                   
                 3 
                 1 
                 Tx 
                 Rx 
                 Rx 
                 Rx 
               
               
                   
                 1 
                 2 
                 Rx 
                 Rx 
                 Rx 
                 Rx 
               
               
                   
                 2 
                 2 
                 Tx 
                 Tx 
                 Rx 
                 Rx 
               
               
                   
                 3 
                 2 
                 Rx 
                 Tx 
                 Tx 
                 Tx 
               
               
                   
                   
               
             
          
         
       
     
         [0102]    As can be seen, the associated chips  602  and  604  channels correspond, such that when a channel on one chip is transmitting or receiving, the corresponding channel on the other chip is doing the opposite. 
         [0103]    Referring now to  FIG. 15   a , there is illustrated the RF isolation link  600  within a chip package. As discussed previously in  FIG. 15 , there are illustrated chips  1602   a  and  1602   b  interconnected by four separate channels  604 . Each channel  604  is represented by two bond wires interconnecting the transformers (not shown) within each of chips  1602   a  and  1602   b . Each of chips  1602   a  and  1602   b  are also connected to various bond pads  1504  within the package by a connection line  1542  that provide connections to the other electronic circuitry. 
         [0104]    The embodiment of  FIG. 15   a  is referred to as a Asplit lead-frame@ package. This is facilitated with the use of a lead frame  1550  on one side thereof and a lead frame  1552  on the other side thereof. Lead frame  1550  is interfaced with terminals  1554  and lead frame  1550  is interfaced with terminals  1556 . During fabrication, the lead frames  1550  and  1556 , which are not electrically connected to each other, provide support for the chips  602   a  and  602   b , respectively. When the chips  602   a  and  602   b  are bonded onto their respective portions of the lead frame, they are then bonded to the appropriate terminals  1554  and  1556  and then the bond wires  604  disposed therebetween. The entire package is then encapsulated in a conventional encapsulate. Thus, the bond wires  604  each comprise a high frequency transmission line disposed between the two chips, each transformer associated with two band wires that provide a Atwo-wire@ transmission line. 
         [0105]    Referring now to  FIG. 15   b , there is illustrated a side view of one of the bond wires  604 . It can be seen that the substrate associated with the die  602   a  has disposed thereon a bonding pad  1560  and the die  602   b  has disposed thereon a bonding pad  1562 . The bond wire  604  is bonded to the pad  1516  on one side with a conventional bond and also to the pad  1562  on the die  602   b . The length of the bond wire  604  is a fraction of a wavelength at the  2 . 4  GHz frequency. However, it will be inductive in nature and will have distributed inductance and capacitance associated therewith. As such, the transmission characteristics of the bond wire can affect the transmission of information between the two dies  602   a  and  602   b . As noted herein above, the input impedance to each of the pads  1560  is on the range of 500 ohms. Thus, for ideal transmission of the information, there might be some matching circuitry required in addition to just the bond wires  604  forming the two-wire transmission line, although that has not been set forth herein. 
         [0106]    Referring now to  FIG. 16 , there is illustrated the manner in which the RF isolation link  600  represented as RF isolation circuitry  1602  may be integrated into two separate multi-functional dies,  1604  and  1606 , within a single package  1608 . The RF isolation circuitry  1602  may provide isolation between components on two separate dies  1604  and  1606 . Associated with one or both of the dies could be additional circuitry  1610  such as a microcontroller or other electronic component. This additional circuitry would be isolated from components in the other die via the RF isolation link  1602 . 
         [0107]    Referring now also to  FIG. 16   a , when an RF isolation link  600  is integrated onto two separate dies  1604  and  1606  in a single package  1608 . The isolation interface  1602 , which includes the transceivers  1612  and the transformers  1614 , may be used to provide simply a digital IN, digital OUT package  1608 . In this embodiment, the digital input  1620  is applied to a first transceiver  1612   a . Alternatively, the digital input  1620  could be applied to digital circuitry connected to the transceiver  1612   a . The isolation circuit operates in the manner described herein above and a second digital output  1622  is provided from transceiver  1612   b  or associated digital circuitry. 
         [0108]    Referring now to  FIG. 16   b , rather than providing a solely digital input/digital output circuit, a single package  1608 , including first and second dies  1604 ,  1606  implementing the RF isolation circuit described herein above, may provide a circuit with a digital input/output and an analog input/output. In this case, a digital input/output  1924  would connect with transceiver  1612   a  or digital circuitry of a first die  1604 . The first die  1604  is coupled with the second die  1606  via the described RF isolation link, and the transceiver  1612   b  is coupled to an analog input/analog output  1626  through a data converter, either an ADC  1614  or a DAC  1616 , depending upon the direction. 
         [0109]    Referring now to  FIG. 16   c , a single package  1908 , including first and second dies  1604 ,  1606  implementing the RF isolation circuit described herein above, may provide a circuit with an analog input/output and on one side and an analog input/output on the other side. In this case, an analog input/output  1640  would connect to an A-D converter  1642  and a D-A converter  1644  and then to the transceiver  1612   a  or digital circuitry of a first die  1604 . The first die  1604  is coupled with the second die  1606  via the described RF isolation link, and the transceiver  1612   b  is coupled to an analog input/output  1646  via an A-D converter  1648  and D-A converter  1650 . In this way, analog signals may be transmitted in either direction across the single package  1608 . 
         [0110]    Referring now to  FIG. 17   a , there is illustrated a chip  1702  including a portion of the RF isolation link described herein above. The chip  2002  includes a single transformer  1704  and the transmit and receive circuitry  1706  of the RF isolation link  600 . The RF isolation link  600  consisting of the transceiver  1706  and the transformer  1704  is integrated with a microcontroller unit  1708  through a digital input/output  1710 . A memory  1712  stores operating instructions and data needed by the microcontroller unit  1708 . The chip  1702  would be able to interconnect with a second chip that included an interface consisting of a transformer  1704  and transceiver  1706  similar to that included within the chip  1702 . By interconnecting to such chips, the microcontroller  1708  and the interconnected chip would be voltage isolated from each other via the complete RF isolation link between them. 
         [0111]    The transmit and receive circuitry  1706  is part of the I/O interface for the integrated circuit. One type of integrated circuit that provides the overall functionality of that illustrated in  FIG. 17   a  is a conventional microcontroller unit of the type C8051FXXX, manufactured by the present Assignee. This chip provides onboard processing through the MCU  1708 , interface to the analog domain and interface to the digital domain. This integrated circuit also has the ability to configure the various outputs and, as such, a digital output could be provided on a serial interface for driving the transmit/receive circuitry  1706  or receiving the serial data therefrom. 
         [0112]    The process of fabricating the MCU  1708 , memory  2012  and the digital I/O  1710 , in addition to the various analog-to-digital data converters or digital-to-analog data converters is fairly complex. As such, the transmit and receive circuitry  1706  and the transformer  1704  must be compatible with the process rather than making the process compatible with the transformer. As will be described herein below, there are a plurality of metal layers utilized to fabricate various interconnects associated with fabrication of the integrated circuit. By utilizing the various metal layers that are already present in the fabrication process, the two sides of the transformer  1704  can be fabricated and isolated from one another with sufficient isolation to provide adequate over voltage protection. Additionally, due to the high voltages and the high frequency of the transformer, the transformer  1704  is actually disposed in a separate portion of the chip surface area such that it does not overlie any of the circuitry associated with the digital operation or the analog operation, since this is a mixed-signal integrated circuit. 
         [0113]    One example of this is illustrated in  FIG. 17   b , wherein the chip  1702  including an RF isolation link consisting of transformer  1704  and transceiver  1706  is integrated with a microcontroller unit  1708  through a digital input/output  1710 . The MCU  1708  also includes an associated memory  1712 . In this case, the first portion of the RF isolation link consisting of a transformer  1704  and transceiver  1706  is interconnected with a second portion of the RF isolation link consisting of transformer  1714  and transceiver  1716 . In this case, the chip  1718  including the second portion of the RF isolation link includes a digital-to-analog converter  1720  and an analog-to-digital converter  1722  for converting the digital output of the transceiver  1716  of the RF isolation link into an analog output and for converting received analog inputs into digital inputs. The chip  1718  enables both the output of an analog signal at analog output  1724  and the input of analog signals at analog input  1726 . These analog signals may then be used in any desired fashion by a circuit designer. 
         [0114]    Referring now to  FIGS. 18   a ,  18   b ,  19  and  20 , there is illustrated the structure of the transformer coils of transformer  714  or  718  ( FIG. 7 ) integrally formed on a CMOS device. Each transformer  714  and  718  is integrated as a part of one of the chips or dies including the RF isolation link. Referring more particularly to  FIGS. 18   a  and  18   b , there are illustrated the two coils included in each of transformers  714  or  718 . A first coil  1802  consists of a first terminal  1804  and a second terminal  1806  formed in the metal layer of a chip referred to as the AMetal 1@ layer. Each of the terminals in the Metal 1 layer are connected to the transformer coil  1808  which resides on a second metal layer of a chip referred to as the AMetal 2@ layer. A conductive via  1810  interconnects the coil  1808  with terminal  1804 . A second connective via  1812  interconnects the coil  1808  with the second terminal  1806 . A second coil resides upon a fifth metal layer referred to as the AMetal 5@ layer. This coil consists of a first bonding pad  1814  and a second bonding pad  1816 . Each of the first and second conductive pads  1814 ,  1816  are interconnected by a second coil  1818  encircling pad  1816  and interconnecting with pad  1814 . Unlike the coil described in  FIG. 18   a , coil  1818  includes both the bonding pads  1814 ,  1816  and the coil  1818  on the same metal layer (Metal 5). 
         [0115]    Typically, the Metal 5 layer is the uppermost layer. Referring now also to  FIG. 19 , there is illustrated the overlapping view of the first and second coils of a transformer on a chip. It can be seen that the pad  1816  is dimensioned such that it is 1/32μx94 μn. The entire coil is dimensioned to be 268 μm by 205 μm. The pad  1814  is dimensioned to the 70 μm×80 μm. The two coils  1818  and  1808  are similar in their configuration and are oriented such that they are substantially Anon-overlapping. @ However, they could overlap. 
         [0116]    Referring now to  FIG. 20 , there is illustrated a side view of a chip  602  containing a transformer structure as described with respect to  FIGS. 18   a ,  18   b  and  19 . The chip  602  includes a substrate layer  2002  containing the transceiver circuitry of the RF isolation link and any electronic circuitry integrated with the RF isolation link as discussed previously. The Metal 1 layer  2004  resides upon the substrate  2002  and includes the first and second terminals  1804 ,  1806  of the first transformer coil. On top of the Metal 1 layer is the Metal 2 layer  2006  containing the first coil  1808  interconnected by vias to the first and second terminals  1804  and  1806  (not shown). Finally, the Metal 5 layer resides over the Metal 2 layer  2008 . The Metal 5 layer  2010  contains the other portion of the transformer, including the bond conduction pads  1816  and the bond pad  1814  (not shown) and the coil  1818  interconnecting the bond pad  1816  with the bond pad  1814 . The Metal 1 layer for the transformer is utilized primarily to provide interconnects to the remaining of the circuits for the terminals  1804  and  1806 . However, the process utilizes all five metal layers for the various interconnects. For the purposes of over voltage protection, it is desirable to separate the coil  1818  from the coil  1808  by as much distance as possible; realizing that the material disposed therebetween is silicon dioxide, a dielectric. An additional concern is the capacitor loading on the coil  1818  to ground, the substrate  2002  typically being disposed at ground. The high voltage will be present on the coil  1818  and, therefore, it is separated from both the substrate and the coil  1818  by as much distance as possible. Although the coil  1818  could have been fabricated in the Metal 1 layer, there would then have been a requirement to provide an interconnection from the ends of the coil to the circuitry. This would have required a Arun@ to be provided beneath the Metal 1 layer, which would require utilization of a polycrystalline layer. Even siliciding of the poly layer would not provide as good a conductive layer as that associated with a metal layer. As such, the configuration utilizes the Metal 1 layer for the interconnects and the Metal 2 layer for the coil. 
         [0117]    Although it would be desirable to provide an even additional metal layer to further separate the coil  1818  from the coil  1808 , it is not feasible to complicate a process with a special additional layer. The only reason that an additional layer would be utilized would be for the purpose of fabricating other circuitry on the integrated circuit. The reason for this is that, once a process is defined as being able to utilize multiple metal layers, substantially all circuits run through that process will use the multiple layers. It would be difficult to dedicate a process for a single integrated circuit that only used that additional metal layer and, therefore, the coil is fabricated from already existing metal layers in an existing process. However, if an additional metal layer were utilized in an existing process in the future, then it is possible that the coil  1818  would be disposed in an even higher layer than Metal 5. 
         [0118]    Referring now to  FIG. 21 , there is illustrated the offset used between metal runs  2102  of the coil  1818  on the Metal 5 layer and metal runs  2104  of the coil  1808  on the Metal 2 layer. Rather than having metal runs  2104  on the Metal 2 layer disposed directly below a metal run  2102  on the Metal 5 layer, they are offset diagonally from each other in order to increase the breakdown voltage between the components by increasing the distance. In the disclosed embodiment, the total distance between the Metal 5 layer run  2102  and the Metal 2 layer run  2404  is 3.63 μm. The Metal 2 layer run  2104  is vertically displaced from the Metal 5 layer run  2102  by 3.54 μms and horizontally displaced by 0.8 μm. The Metal 5 run layer  2102  is vertically separated from the silicon layer by 5.24 μm. This structure should provide a breakdown voltage between the Metal 5 and Metal 2 layers according to the equation 3.63×10 −6  m*8×10 8  v/m=2904 v of breakdown voltage isolation. The breakdown voltage between the Metal 5 layer  2402  and the silicon layer  2406  can be determined according to the equation 5.24×10 −6  m*8×10 8  v/m=4192 v. 
         [0119]    Referring now to  FIG. 22 , there is illustrated a cutaway perspective view of the coils  1818  and  1808  illustrated in  FIG. 21 . It can be seen that the metal runs  2104  are substantially the same shape as the metal runs  2102  but they are non-overlapping and separated by a dielectric layer. This illustration illustrates only a single corner of the coils. 
         [0120]    Referring now to  FIG. 23 , there is illustrated a chip  602  including an RF isolation link according to the present disclosure. The area of the chip  602  would be divided into at least two sections. A first section  2302  would contain the circuitry for providing the transformer for electromagnetically coupling with a transformer on another chip to provide the voltage isolation link between the chips. The remaining electronic circuitry of the chip would be located in a separate area  2304  and would include the transmitter and receiver circuitry of the voltage isolation link associated with the transformer as well as any electronic circuitry that would be integrated with the voltage isolation link, such as a microcontroller or other type of electronic device. This would be repeated for multiple voltage isolation links for additional data paths. Additionally, it is noted that the layout is such that the area  2302  that contains the transformer on the upper surface thereof will have provided the pads  2116  in the center of the coil  2118  and the pad  2114  on the exterior thereof. The pad  2114  is located proximate the edge of the chip such that the bond wire  604  can be bonded thereto. Additionally, the pad  2116  is on the same surface as the pad  2114  such that the bond wire  604  associated therewith can be connected thereto. As such, there are no runs required to connect to the pad  2116  in a coil that would be required to run through other layers and run closer to the coils therein at right angles thereto. The bond wire  604  associated therewith will actually be disposed farther away from the actual metal runs  2102  associated with the coil  1818 . An additional area could be included on the chip for additional electronic circuitry to be voltage isolated via a voltage isolation link on the same chip. 
         [0121]      FIG. 24  illustrates the overall structure of the RF isolation link implemented on a chip  2402 . Four separate interface connections  2404  provide connection of each of the four channels of the RF isolation link integrated into the chip  2402 . Each of the four interfaces  2404  is linked with the oscillator  2406  and coil  2408 . Connected to each of the interfaces  2404  are the transformers  2410  consisting of a first coil  2412  and a second coil  2414 . Coil  2414  connects with the interface  2404  to provide interconnection with an external chip via the RF isolation link. Coil  2412  interconnects to bond pads  2416 . It is noted that the channel one and channel four coils  2414  each include two separate bond pads  2416 . However, the channel two and three coils  2414  each have a bond pad within the interior of the coil but share the external bond pad  2416 x between channels two and three. Pad circuitry  2418  is associated with the oscillator circuit  2406  and the coils  2410 . The pad circuitry  2418  is interconnected with the remainder of the circuitry on a chip  2402  via a number of bond pads. The bond pads comprise a ground bond pad  2418 , a V DD  bond pad  2420 , two enable bond pads  2422 , four output bond pads  2424  and four input bond pads  2426 , one for each channel. 
         [0122]    One problem with the above-described RF isolation link design is that RF interference from nearby transmitting cellular telephones may create common mode interference that may not be filtered in the receiving portion. Referring now to  FIG. 25 , at GHz frequencies the application printed circuit board consisting of two separate portions  2502  creates split ground planes which may act as a dipole antenna. The split ground planes may have dimensions that are close to the quarter wavelength dimension at 900 MHz. This results in very large common mode signals which may be passed through the isolator chip  2504 . Measurements from a nearby transmitting GSM cell phone at maximum power can create common mode voltages of as high as 3.4V peak at 900 MHz. This would cause interference within the RF isolation link as described herein above, causing a A0@ to be incorrectly detected as a A1@ when a cell phone was operating nearby. One manner for reducing this problem is by adding an EMI capacitor  2506  between the isolated ground planes. Thus, at a frequency of 900 MHz, a circuit without the EMI capacitor  2506  would have a 3.4V peak common mode voltage but with a 300 pF capacitor  2506  would only have a 1.1V peak. Likewise, at 2 GHz, the circuit without an EMI capacitor  2506  would have a 0.85V peak common mode voltage and a 0.07V peak common mode voltage when a 300 picofarad EMI capacitor  2506  was included. An RF isolator as described herein above cannot handle this level of common mode interference. 
         [0123]    The previously described single-ended design relies upon the transformer to provide all common mode rejection. While the transformer has very good common mode rejection below 100 MGz, the common mode rejection for the transformer is poor at GHz frequencies. This is due to the parasitic capacitances  2602  that are created within the transformer  2604  as illustrated in  FIG. 26 . This is more fully illustrated in  FIG. 27  wherein the vertical axis illustrates the common mode gain and the horizontal axis illustrates frequencies. As can be seen in  FIG. 27 , at 100 MHz frequencies, the common mode gain is relatively minimal. However, as the GHz frequencies are approached, the common mode gain increases, thus increasing the amount of common mode interference which would be passed through the transformer circuit of the RF isolation link. 
         [0124]    In the embodiment illustrated in  FIG. 28 , the problems of common mode interference are addressed by modifying the transformer  2802  to be a center tapped transformer and including a differential amplifier  2810 . Use of the center tapped transformer  2802  moves out the frequency at which the circuit resonates by splitting the parasitic capacitances. The center tap of transformer  2802  on the transmitter side is connected through a capacitor  2804  to ground. The center tap of transformer  2802  on the receive side is grounded. The bandwidth of the center tap transformer is two times higher than the single ended design for common mode signals. This helps suppress  900  MHz common mode interference. The data to be transmitted is applied to a first input of NAND gate  2814  and the RF signal is applied to second input of NAND gate  2814  before being applied to the center tapped transformer  2802 . A differential amplifier  2810  is used on the receive side to further suppress common mode interference. In this circuit, common mode interference is applied to the inputs of the differential amplifier  2810  as a common mode signal which is rejected by the differential amplifier  2810 . The transmitted RF signal is differential and is gained up by the receiver RF amplification and applied to a detector circuit  2810 , one example of which may be the detector circuits described herein above. 
         [0125]    Referring now to FIGURE.  29 , there is illustrated an alternative embodiment of the RF isolation link  2900  of  FIG. 28  consisting of transmitter circuitry  2902  and receiver circuitry  2904 . The transmitter circuitry  2902  consists of a NAND gate  2908  having a first input connected to receive the data to be transmitted over the RF isolation link  2900  and a second input connected to receive the RF carrier signal. The RF carrier in the preferred embodiment comprises a 2 GHz signal. The data input to the first input of the NAND gate  2908  consists either of a logical A1@ or A0@ which will selectively gate the RF carrier signal to the output of NAND gate  2908  in the presence of a logical A1.@ This causes the output of the NAND gate to either provide the RF carrier signal when the data bit is A1@ or not provide the RF signal when the data bit is A0.@ The output of the NAND gate  2908  is connected to the gate of a p-channel transistor  2910 . The drain-source paths of the p-channel transistor  2910  are connected between V DD  and a first input of transformer  2912 . The transformer  2912  is a center tap transformer having its center tap node  2914  connected to a transistor  2916 . The drain-source path of transistor  2916  is connected between node  2914  and ground. The gate of transistor  2916  is connected to receive signal tx_ena-bar. The output of NAND gate  2908  is also connected to an input of inverter  2918 . The output of inverter  2918  is connected to the gate of transistor  2920 . The drain-source path of transistor  2920  is connected between transformer  2912  and ground. A receiver amplifier  2922  is connected across transformer  2912  and may be disabled by a disable input  2924  when the chip is transmitting. The transformer  2912  electromagnetically couples the RF carrier signal to transformer  2926  via bond wires  2928 . This links the data represented by the RF carrier signal between the transformers and limits common mode signals while providing voltage isolation between the chips via the first and second transformers  2912  and  2926 . Each of the transformers  2912  and  2926  are associated with opposite sides of the interface. 
         [0126]    The receiver circuitry  2904  receives the signal which has been electromagnetically coupled via the center tap transformer  2912  onto the bond wires  2928  to center tap transformer  2926 . Connected to a center tap node  2930  of center tap transformer  2926  is a transistor  2932 . The drain-source path of the transistor  2932  is connected between center tap node  2930  and ground. The gate of transistor  2932  is connected to V DD . The outputs of center tap transformer  2926  are connected to the inputs of a differential amplifier  2934 . The differential amplifier  2934  consists of a first stage  2936  and second stage  2938  providing common mode rejection and a third stage  2940  providing single ended gain. 
         [0127]    The first stage  2936  consists of a set of two p-channel transistors  2942 ,  2944 , and two n-channel transistors  2946  and  2948 . The drain-source path of transistor  2946  is connected between node  2950  and node  2952  connected to center tap transformer  2926 . The gates of transistors  2946  and  2948  are cross coupled through capacitors  2956  and  2958  to nodes  2950  and  2956 , respectively. Transistor  2942  has its drain-source path connected between V DD  and node  2952 . Transistor  2948  has its drain-source path connected between node  2954  and node  2956 . Transistor  2944  has its drain-source path connected between node V DD  and node  2954 . The gate of transistor  2942  is connected to node  2952 . The gate of transistor  2944  is connected to node  2954 . A resistor  2962  is additionally connected between the gate of transistor  2946  and a bias node  2964 . A resistor  2966  is also connected between the gate of transistor  2948  and the bias node  2964 . 
         [0128]    The second stage  2938  is connected to the first stage  2936  at nodes  2952  and  2954 . Transistor  2968  has its gate connected to node  2952 . Transistor  2970  has its gate connected to node  2954 . The drain-source path of transistor  2968  is connected between node  2972  and node  2974 . Transistor  2970  has its drain-source path connected between node  2976  and node  2974 . A current source  2978  is connected between node  2974  and ground. Transistor  2980  has its drain-source path connected between node  2972 . The gate of transistor  2980  is connected to node  2972 . Transistor  2982  has its drain-source path connected between V DD  and node  2976 . The gate of transistor  2982  is connected to node  2972 . Transistor  2984  has its gate connected to node  2976 . The drain-source path of transistor  2984  is connected between V DD  and node  2976 . A current source  2986  is connected between node  2976  and ground. 
         [0129]    The third stage  2940  connects with the second stage  2938  at node  2976 . A capacitor  2988  is connected between node  2976  and an input of amplifier  2990 . The output of amplifier  2990  has a feedback resistor  2992  connected to its input. The output of amplifier  2990  is also connected to a detector circuit  2994  for detecting the amplified data coming from the gained amplifier. A transmitter circuit  2926  connects to the single tap transformer  2926  at node  2950 . The transistor  2928  has its drain-source path connected between node  2956  and ground. The gate of transistor  2928  is also connected to ground. 
         [0130]    Referring now to  FIGS. 30   a ,  30   b ,  31  and  32 , there is illustrated the structure of the transformer coils of transformer  2912  or  2926  ( FIG. 7 ) integrally formed on a CMOS device. Each transformer  2912  and  2926  is integrated as a part of one of the chips or dies including the RF isolation link. Referring more particularly to  FIGS. 30   a  and  30   b , there are illustrated the two coils included in each of transformers  2912  or  2926 . A first coil  3002  consists of a first terminal  3004  and a second terminal  3006  formed in the metal layer of a chip referred to as the AMetal 1@ layer. Each of the terminals in the Metal 1 layer are connected to the transformer coil  3008  which resides on a second metal layer of a chip referred to as the AMetal 2@ layer. A conductive via  3010  interconnects the coil  3008  with terminal  3004 . A second connective via  3012  interconnects the coil  3008  with the second terminal  3006 . A second coil resides upon a fifth metal layer referred to as the AMetal 5@ layer. This coil consists of a first bonding pad  3014  and a second bonding pad  3016 . Each of the first and second conductive pads  3014 ,  3016  are interconnected by a second coil  3018  encircling pad  3016  and interconnecting with pad  3014 . Unlike the coil described in  FIG. 30   a , coil  3018  includes both the bonding pads  3014 ,  3016  and the coil  3018  on the same metal layer (Metal 5). 
         [0131]    Typically, the Metal 5 layer is the uppermost layer. Referring now also to  FIG. 31 , there is illustrated the overlapping view of the first and second coils of a transformer on a chip. It can be seen that the pad  3016  is dimensioned such that it is 70 μm×70 μm. The entire coil is dimensioned to be 205 μm by 205 μm. The pad  3014  is dimensioned to the 70 μm×70 μm. The two coils  3018  and  3008  are similar in their configuration and are oriented such that they are substantially Anon-overlapping. @ However, they could overlap. The center tap is provided on the M1 layer with a strip  3104  extending all the way across coils of the transformer and including a conductive via  3102  providing the center tap interconnecting the Metal 1 layer to the Metal 2 layer in coil  3008 . 
         [0132]    Referring now to  FIG. 32 , there is illustrated a side view of a chip  3200  containing a transformer structure as described with respect to  FIGS. 30   a ,  30   b  and  31 . The chip  3200  includes a substrate layer  3202  containing the transceiver circuitry of the RF isolation link and any electronic circuitry integrated with the RF isolation link as discussed previously. The Metal 1 layer  3204  resides upon the substrate  3202  and includes the first and second terminals  3004  and  3006  of the first transformer coil. On top of the Metal 1 layer is the Metal 2 layer  3206  containing the first coil  3008  interconnected by vias to the first and second terminals  3004  and  3006  (not shown). Finally, the Metal 5 layer resides over the Metal 2 layer  3008 . The Metal 5 layer  3210  contains the other portion of the transformer, including the bond conduction pads  3016  and the bond pad  3014  (not shown) and the coil  3018  interconnecting the bond pad  3016  with the bond pad  3014 . The Metal 1 layer for the transformer is utilized primarily to provide interconnects to the remaining circuits for the terminals  3004  and  3006 . However, the process utilizes all five metal layers for the various interconnects. For the purposes of over voltage protection, it is desirable to separate the coil  3018  from the coil  3008  by as much distance as possible; realizing that the material disposed therebetween is silicon dioxide, a dielectric. An additional concern is the capacitor loading on the coil  3018  to ground, the substrate  3202  typically being disposed at ground. The high voltage will be present on the coil  3018  and, therefore, it is separated from both the substrate and the coil  3018  by as much distance as possible. Although the coil  3018  could have been fabricated in the Metal 1 layer, there would then have been a requirement to provide an interconnection from the ends of the coil to the circuitry. This would have required a Arun@ to be provided beneath the Metal 1 layer, which would required utilization of a polycrystalline layer. Even siliciding of the poly layer would not provide as good a conductive layer as that associated with a metal layer. As such, the configuration utilizes the Metal 1 layer for the interconnects and the Metal 2 layer for the coil. The center tap strip  3104  runs through the Metal 1 layer and connects to the coil  3008  in the Metal 2 layer using conductive via  3102 . 
         [0133]    Although it would be desirable to provide an even additional metal layer to further separate the coil  3018  from the coil  3008 , it is not feasible to complicate a process with a special additional layer. The only reason that an additional layer would be utilized would be for the purpose of fabricating other circuitry on the integrated circuit. The reason for this is that, once a process is defined as being able to utilize multiple metal layers, substantially all circuits run through that process will use the multiple layers. It would be difficult to dedicate a process for a single integrated circuit that only used that additional metal layer and, therefore, the coil is fabricated from already existing metal layers in an existing process. However, if an additional metal layer were utilized in an existing process in the future, then it is possible that the coil  3018  would be disposed in an even higher layer than Metal 5. 
         [0134]    Another concern in reducing common mode rejection is the ability to set the receiver gain and transmit power to a level to reliably pass through data but no higher. This conserves power in the transmitter and improves common mode rejection which is worse at higher receiver gains. Once this gain is established, it should remain constant over temperature and process changes to provide optimal system performance. This can be achieved by setting the power supply voltages (V DD ) to the transmitter and the receiver to vary with temperature and process instead of being a constant regulated voltage. This is illustrated in  FIG. 33 . As can be seen, for both a slow process and fast process, the voltage V DD  increases as the temperature increases. This helps to keep the RF gain of the amplifier more constant as temperature changes and allows lower supply currents at lower temperatures. 
         [0135]    Referring now to  FIG. 34 , there is illustrated a prior art method for generating the reference voltage wherein the PTAT current generator  3402  is connected to the gate of transistor  3404 . The drain-source path of transistor  3404  is connected between voltage and node  3406 . A resistor  3408  is connected between node  3406  and transistor  3410 . The emitter/collector pathway of transistor  3410  is connected between transistor  3408  and ground. The base of transistor  3410  is connected to its collector. 
         [0136]      FIG. 35  illustrates the modified method for generating the band gap reference voltage such that the voltage will vary with respect to temperature. The PTAT current generator  3402  again provides a voltage to the gate of transistor  3404  which provides a PTAT current. The PTAT current provided by the PTAT current generator  3402  is proportional to absolute temperature. The source-drain pathway of transistor  3404  is connected between voltage and node  3406 . A p-channel transistor  3502  has its source-drain pathway connected between node  3406  and node  3504 . The gate of transistor  3502  is also connected to node  3504 . A resistance  3506 , which is larger than the resistance of resistor  3408  in  FIG. 34 , is connected between node  3504  and ground. By setting the size of the PMOS transistor  3502  and the resistance  3504 , the reference voltage can be set to a desired level. Since the bias current provided to the receiver is a PTAT current, this keeps the receiver gain constant. 
         [0137]    Referring now to  FIG. 36 , in switching power supplies, there is a need for gate drivers which drive the power MOSFETs or IGBTs connected to the power transformer. Drivers on the secondary side are typically controlled by a PWM controller on the primary side, and thus, the connection to the drivers from the PWM controller requires high voltage isolation. The power transformer  3602  includes a primary side  3604  and a secondary side  3606 . Connected to each end of the primary side  3604  of the power transformer  3602  is a pair of power transistors  3608 . The drain/source path of transistor  3608   a  is connected between the input voltage (V IN ) and node  3610 . The drain/source path of transistor  3608   b  is connected between node  3610  and ground. The drain/source path of transistor  3608   c  is connected between V IN  and node  3612 . The drain/source path of transistor  3608   d  is connected between node  3612  and ground. The gate of each transistor  3608  is connected to a driver  3614  that is connected to the PWM controller  3616 . 
         [0138]    The PWM controller  3616  provides switching signals to the power transistors  3608  which are turned on and off responsive to the switching signals provided to the drivers  3614 . The PWM controller  3616  also provides switching signals to transistors  3618  on the secondary side  3606  of power transformer  3602  through the isolation barrier  3620 . The drain/source path of transistor  3618   a  is connected between node  3622  and ground. The drain/source path of transistor  3618   b  is connected between node  3624  and ground. The gates of transistors  3618  are connected to drivers  3626  which receive signals from the PWM controller  3616  through the isolation barrier  3620 . Each end of the secondary side  3606  of the power transformer  3602  is connected between nodes  3624  and node  3622 . An inductor  3628  is connected between node  3624  and V OUT . An inductor  3630  is connected between node  3622  and V OUT . Finally, a capacitor  3632  is connected between V OUT  and ground. Thus, there must be some means for voltage isolating the signals provided over the isolation barrier  3620  to the secondary side transistors  3618  from the PWM controller  3616 . 
         [0139]    Currently, this problem is solved in a number of non-integrated fashions. A first common method, illustrated in  FIG. 37 , makes use of opto-isolators. In this solution, the PWM controller  3702  provides the control signals through a resistor  3704  to the base of a transistor  3706 . The emitter/collector pathway of the transistor  3706  is connected between the optical isolator  3708  and ground. The optical isolator  3708  is connected to V DD  through a transistor  3710 . The optical isolator  3708  consists of a light emitting diode  3712  between resistor  3710  and the emitter of transistor  3706  and a light detecting transistor  3714 . The emitter of transistor  3714  is connected to V DD  through a resistor  3716 . The collector of transistor  3714  is connected to ground. The emitter of transistor  3714  is also connected to the gate driver integrated circuit  3718  which provides a signal to the power FET  3720 . 
         [0140]    An alternative prior art solution uses a pulse transformer as illustrated in  FIG. 38 . The PWM controller  3802  provides control signals to a driver  3804 . The driver  3804  provides pulses which are transmitted electromagnetically through a transformer  3806 . The pulses are received at a receiver  3808  and used to operate a gate driver  3810 . 
         [0141]    A third prior art alternative, illustrated in  FIG. 39 , uses an integrated digital isolator  3904  with a separate gate driver IC. In this case the PWM controller  3902  connects to the digital isolator  3904  which connects to the driver IC  3906 . The digital isolator  3904  and the gate driver IC  3906  provide isolation between the PWM controller  3902  and the power FET  3908  connected to the driver IC  3906 . This method is currently the fastest system and is smaller than other implementations. However, this implementation is expensive due to the high cost of the digital isolator  3904 . 
         [0142]    Referring now to  FIG. 40 , there is illustrated the implementation of an embodiment wherein an isolated gate driver  4002  is used to voltage isolate the PWM controller  4004  from the power FET circuitry  4006 . The isolated gate driver  4002  combines a digital isolator with a gate driver into a fast, integrated, low cost isolated gate driver. This provides a few substantial benefits to the isolation circuitry. First, the cost is substantially less since only a single IC is necessary to provide isolation rather than the two chips discussed in  FIG. 39 . Furthermore, the single isolated gate driver IC will have a lower delay than the implementation discussed in  FIG. 39  since the digital isolator  3904  of  FIG. 39  uses a substantial part of its delay in driving signals off of the digital isolator chip  3904 . This requirement is eliminated in the integrated solution wherein the isolator and gate driver are on the same chip. 
         [0143]    The general structure of the integrated isolator and gate driver of the present disclosure is illustrated in  FIG. 41 . This structure includes the isolation structures described herein above and further including a gate driver with said isolation structure. The isolated gate driver includes a NAND gate  4102 . The NAND gate  4102  is connected to receive the data to be transmitted through the isolation link. In this case, the data comprises the control signals from the PWM controller. The NAND gate  4102  is additionally connected to receive an RF signal. The RF output of the NAND gate  4102  is connected to the input of an inverter  4104 . The output of the inverter  4104  is connected to a first transformer  4106 . The transformer  4106  electromagnetically couples the provided PWM controller signals to a second transformer  4108 . The output of the second transformer  4108  is connected to a receiver and detector circuit  4110  which may be configured in any of the manners discussed herein above. The output of the receiver and detector circuit  4110  is provided to the input of an inverter amplifier  4112  which is connected to the gate driver  4114  that drives a connected power transistor. 
         [0144]    Referring now to  FIG. 42 , there are illustrated the two separate dies  4202  and  4204  integrated upon a single package  4206  providing the integrated digital isolator and gate driver. In previous embodiments of the digital isolator, die one  4202  and die two  4204  are implemented in 0.25 μm CMOS technology. The 0.25 μm CMOS technology is needed to process the 2.1 GHz RF carrier signal provided at the NAND gate  4102  of  FIG. 41 . However, power MOSFET gate driver ICs typically have to drive between 10 V and 20 V. High voltage transistors capable of supporting these voltage ranges are not available in the 0.25 μm CMOS process. Thus, an 18 V CMOS process with high voltage NMOS and PMOS transistors that provides 0.35 μm, 3.3 V CMOS logic transistors must be used in implementing the circuitry within dies  4202  and  4208 . With this process, it is possible to integrate the 10-20 V gate driver using the high voltage transistors operating at an 18 V range, and the RF receiver using the 0.35 μm logic transistors operating at a 3.3 V range. 
         [0145]    Referring now to  FIG. 43 , there is provided a more detailed illustration of the circuitry for implementing the isolated gate driver IC. As described previously, the NAND gate  4302  is connected to receive the data stream from the PWM controller and the RF carrier signal. The output of the NAND gate  4302  is connected to the gate of transistor  4302  and the input of an inverter  4304 . The output of inverter  4304  is connected to gate of transistor  4306 . The drain/source path of the transistor  4306  is connected between transformer  4306  and ground. The source/drain path of transistor  4302  is connected between 3.3 V and transformer  4306 . 
         [0146]    The transformer  4310  of the isolation link is a center tap transformer. The outputs of the transformer  4310  are connected to separate inputs of a differential amplifier circuit  4312 . The output of the differential amplifier circuit  4312  is connected to a capacitor  4314 . The other side of the capacitor  4314  is connected to a parallel connection of an inverter  4316  and a resistor  4318 . The other side of the parallel connection of the inverter  4316  and resistor  4318  is connected to another capacitor  4320 . The capacitor  4320  is also connected to a detector circuit  4322  which detects the PWM control signal provided by the PWM controller over the isolation link. A regulator  4324  is connected between the 18 V power source and the detector  4322 . The circuitry between the comparator circuit  4312  up to and including the detector circuit  4322  operate on a 3.3 V supply. The remaining circuitry operates using an 18 V power supply and includes the level shift circuitry  4326  having an input connected to the output of the detector circuit  4322  and an output connected to the driver  4314 . The level shift circuit  4326  increases the voltage level of the detected PWM control signal to a voltage level able to operate the driver  3914 . The output of the driver  3914  would then be connected to the power FET transistors. 
         [0147]    Referring now to  FIG. 44 , there is illustrated a more detailed description of the level shifter circuit  4326 . The input to the level shifter  4326  provided from the detector  4322  is connected to a first inverter  4402 . The output of inverter  4402  is connected to the input of a second inverter  4404  and the gate of a transistor  4406 . The output of inverter  4404  connects to the gate of transistor  4408 . The source/drain path of transistor  4408  is connected between node  4410  and ground. A transistor  4412  has its source/drain path connected between 18 V system power and node  4410 . The gate of transistor  4412  is connected to node  4414 . Also having its gate connected to node  4414  is a transistor  4416 . The source/drain path of transistor  4416  is connected between 18 V system power and node  4414 . A 50 μA current source  4418  is connected between node  4414  and ground. A transistor  4420  has its source/drain path connected between  18  V system power and node  4422 . The gate of transistor  4420  is connected to node  4410 . Transistor  4424  has its source/drain path connected between node  4422  and ground. The gate of transistor  4424  is connected to node  4410 . A transistor  4430  has its source/drain path connected between 18 V system power and node  4410 . The gate of transistor  4430  is connected to the drain of transistor  4432  at node  4434 . The source/drain path of transistor  4432  is connected between 18 V system power and node  4434 . The gate of transistor  4432  is connected to node  4414 . Transistor  4406  has its source/drain path connected between node  4434  and ground. A series connection of inverters  4440  has an input connected to node  4422  and the output thereof would be connected to the driver  3914 . 
         [0148]    Referring now to back to  FIG. 40 , in addition to providing PWM control signals to the drivers on the opposite side of the isolation barrier  4020 , voltage sensing signals indicating the output voltage V out  must be provided from V out  back to the PWM controller  4016  over the isolation barrier  4020 . Since the output voltage is located on the secondary side and the PWM controller  401   b  is located on the primary side, high voltage isolation is again required. The output voltage must be accurately measured (typically with less than a 1% error) and sent as a feedback signal across the isolation barrier  4020 . 
         [0149]    The most common prior art method of isolating the feedback signal provided to the PWM controller  4016  is illustrated in  FIG. 45 . This method employs an opto-isolator  4502 . A voltage divider circuit consisting of resistor  4504  connected to V out  and node  4506  and a second resistor  4508  connected between node  4506  and ground is connected to a first input of an op-amp  4510 . A second input of the op-amp  4510  is connected to a reference voltage generator  4512  that generates a voltage V REF . The op-amp  4510 , based upon the comparison, generates an error voltage V E  which is applied to the input of a driver  4514 . The output of the driver  4514  is connected to the optical isolator  4502  consisting of a light emitting diode  4516  and a light detecting transistor  4518 . The output of the optical isolator  4502  is connected to a detector circuit  4520  that provides the feedback voltage V FB  to the PWM controller  4016 . The problem with the implementation illustrated in  FIG. 45  is that the analog optical isolator  4502  is generally slow (i.e., delay times of one to ten microseconds) and temperature variations will affect the error signal V E . 
         [0150]    Referring now to  FIG. 46 , there is illustrated an alternative embodiment of a means for isolated voltage sensing. In this solution, the voltage sensing process is voltage isolated by an integrated IC package including two die. The RF digital isolator is used to transfer the data across the isolation barrier. A voltage divider consisting of resistors  4602  and  4604  enable the output voltage to be measured and provided to a first input of an operational amplifier  4606 . The first resistor  4602  is connected between V OUT  and node  4608 . The second resistor  4604  is connected between node  4608  and ground. A capacitor  4610  is connected between node  4608  and the output of operational amplifier  4606 . A second input of the operational amplifier  4606  is connected to a reference voltage generator  4612 . 
         [0151]    The reference voltage generator  4612  is programmed via a digital trim memory. The reference voltage will need to be trimmed to meet the 0.5% accuracy that is necessary for measuring the output voltage. This can be performed at IC test by using a one time programmable (OTP) non-volatile memory. This in a preferred embodiment may be a 32 bit memory available from TSMC. The output of the operational amplifier  4606  provides a voltage error signal V E  which is applied to the input of an A/D converter  4616 . The voltage error signal V E  is used as the voltage feedback signal on the primary side. The output of the A/D converter  4616  is provided as a 6-bit digital output to a transmitter/data encoding circuit  4618  wherein the voltage error signal is encoded and transmitted. The output of the transmit/data encoding circuit  4618  is a single bit serial output which is output over the RF isolation link described herein above. 
         [0152]    A data recovery circuit  4620  receives the data from the RF isolation link and recovers the voltage error signal as described herein above. The signal is provided to a digital to analog converter  4622 . The output of the digital to analog converter  4622  provides the voltage error signal as the voltage feedback signal V FB  that is used by the PWM controller as an indication of the output voltage V out  on the secondary side. The speed and resolution of the analog to digital converter  4616  and digital to analog converter  4622  is a function of the loop band width and the output error requirements. A 10 MHz 6-bit ADC is adequate for up to 1.5 MHz PWM frequencies. However, ADC=s having a lower speed may be used since most loop band widths are much lower. 
         [0153]    Referring now to  FIG. 47 , there is illustrated an integrated chip including two isolated gate drivers and an isolated voltage sensing function. This part would integrate many components in a switching power supply and provide isolation for these functions between the primary side and the secondary side. Signal A_IN and signal B_IN are provided to inputs  4702  and  4704  and are provided at output pins  4706  and  4708  as signals A_DRV and B_DRV. This single integrated chip would receive PWM controller signals at input pins  4702  and  4704  and provide output signals for driving power transistors associated with the switched power supply on the secondary side. The inputs and outputs are voltage isolated from each other according to the system described herein above. Additionally, sensing of the output voltage may be obtained between voltage input pin  4710 , connected to V OUT , and voltage feedback pin V FB    4712 , connected to the PWM controller. The isolation of the voltage sensing function between the primary side and secondary side is performed in the same manner as described herein above. Thus, the integrated device  4700  described with respect to  FIG. 47  would provide isolation for drivers on the primary or secondary side of a switched power supply from the PWM controller and provide isolated voltage sensing from the secondary or the primary side from the PWM controller. 
         [0154]    One issue with an RF isolator, such as that described herein above, is the radiated emissions caused by use of the RF carrier for transmitting data. The FCC specifies that the radiated emissions from a device must be less than 500 μV per meter at 3 meters. The use of a balanced driver circuit can help reduce the level of emissions. However, without shielding and when using a half-wave dipole antenna PCB layout (worst case), the emissions from the RF isolator will be approximately 500 μV per meter per channel. Thus, a four channel RF isolator could have emissions as high as 2 mV per meter which would violate the specifications of the FCC in the worst case scenario. This situation is illustrated in  FIGS. 48   a  and  48   b  wherein when a single RF frequency at 2.1 GHz is used to transmit the data over the RF isolator. The single frequency use causes a 2.1 GHz emission peak to appear in the spectrum emissions for the RF isolator. 
         [0155]    One method for greatly minimizing radiated emissions is to use an RF carrier that changes frequency over time. Thus, rather than transmitting using a single carrier wave at 2.1 GHz, the circuitry used to generate the RF carrier signal is modified such that the oscillator constantly sweeps between, for example, 2.1 GHz and 2.2 GHz. This is more fully illustrated in  FIGS. 49   a  and  49   b .  FIG. 49   a  illustrates how the RF carrier signal sweeps between 2.1 GHz and 2.2 GHz in sixteen steps. Thus, at any particular time, rather than only a single frequency being utilized as the RF carrier, any of the sixteen frequencies may be provided for transmitting the data over the RF isolation link. In this manner, rather than the emission spectrum having a single spike at 2.1 GHz, as illustrated in  FIG. 48   b , an emission spectra such as that illustrated in  FIG. 49   b  is provided, wherein sixteen separate peaks are provided between 2.1 GHz and 2.2 GHz. The average peaks at any one frequency are significantly smaller than that of the emission spectra wherein only a single RF frequency is used. 
         [0156]    Either an analog or a digital sweep may be used. The preferred embodiment uses a digital sweep since it is easier to implement. By using sixteen steps between 2.1 GHz and 2.2 GHz, the emission level of the isolator is reduced by a level of sixteen. Since the FCC looks at a 1 MHz band, the steps from the 2.1 GHz frequency to the 2.2 GHz frequency should be set greater than this. While the present disclosure has described having a sweep between 2.1 GHz and 2.2 GHz, it should of course be realized that the sweep may be between any two frequencies. The number of steps may also be set higher to give further emissions reduction from the isolator. 
         [0157]    Referring now to  FIG. 50 , there is illustrated a block diagram of the circuit for providing the stepped RF carrier signal between 2.1 and 2.2 GHz. A slow ring oscillator  5002  generates a 50-60 MHz oscillating signal that is provided via line  5004  to a divider circuit  5006 . The divider circuit  5006  utilizes the 50-60 MHz signal provided by the slow oscillator  5002  to generate a four-bit control code that is used to drive the RF oscillator circuit  5010 . The control code generated by the divider circuit  5006  is provided over a four line bus  5008  to the RF oscillator circuit  5010 . The control code generated by the divider circuit  5006  may include more than four bits, however, only four bits are provided to the RF isolator  5020  over the four line bus  5008 . The RF isolator circuit utilizes the four bit code to generate the sweep signal between the first and second frequency levels and provides the output sweep signal from an output  5012 . Each of the 16 four-bit codes causes the generation of a different frequency between and including the first and second frequency levels. Using the circuit of  FIG. 50 , the RF carrier frequency will change at a 400-500 KHz rate over sixteen frequencies that are 2-4 MHz apart and repeat at a 50-63 KHz rate. 
         [0158]    The circuit of  FIG. 50  uses a free running slow (60-70 MHz) ring oscillator  5002  to charge the RF carrier. This uses a very low current of approximately 50 μAmps. The slow ring oscillator  5002  is illustrated in  FIG. 52 . The ring oscillator  5002  consists of a plurality of inverters  5202  that are in series connection with each other. A series of five inverters  5202  are interconnected with each other and has a feedback loop connected from node  5204  to the input of inverter  5202   a . Inverter  5206  has its input connected to node  5204  and its output connected to inverter  5208 . The output of inverter  5208  comprises the output of the ring oscillator  5002  which is provided to the divider circuit  5006 . The Vdd for the slow oscillator  5002  is derived from the reference voltage which has a large PTAT component. This keeps the oscillation frequency fairly stable over the process and temperatures. 
         [0159]    Referring now to  FIG. 51 , there is more fully illustrated the RF oscillator circuit  5010 . The inputs of the RF oscillator circuit  5010  are connected to receive the four bit codes from the divide circuit  5006  of  FIG. 50 . The four bit codes are provided to the gates of a first group of transistors  5102  and a second group of transistors  5103  to turn the transistors on and off. Each of the four transistors in group  5102  has its source/drain path connected between a capacitor  5104  and ground. At the other end, each of the capacitors  5104  are connected to a node  5106 . Each of the transistors  5103  has its source/drain path connected between a capacitor  5108  and ground. The other side of each of the capacitors  5108  is connected to node  5110 . An additional capacitor  5112  is connected between node  5106  and ground. A capacitor  5114  is also connected between node  5110  and ground. 
         [0160]    Connected between nodes  5106  and  5110  is an inductor  5116 . A transistor  5118  is connected to the inductor  5116  at node  5110  and has its source/drain path connected between node  5110  and ground. The gate of transistor  5118  is connected to the opposite end of the inductor  5116  at node  5106 . Another transistor  5120  is connected to the inductor  5116  at node  5106 . The transistor  5120  has its source/drain path connected between node  5106  and ground. The gate of transistor  5120  is connected to the opposite end of inductor  5116  at node  5110 . Another transistor  5122  has its source/drain path connected between Vdd and node  5106 . The gate of transistor  5122  is connected to node  5120 . A final transistor  5124  has its source/drain path connected between Vdd and node  5110 . The gate of transistor  5124  is connected to node  5106 . An inverter  5126  is connected between node  5106  and the output node  5012  of the RF oscillator  5010 . Responsive to the control codes applied to the first and second groups of transistors  5102  and  5103 , the RF oscillator  5010  will generate a stepped RF carrier signal at its output  5012  between the first and second selected frequencies based upon values of the inductors and capacitors used within the circuit. 
         [0161]    Referring now to  FIG. 53 , there is illustrated an alternative embodiment for the RF carrier generation circuitry wherein the RF oscillator  5010  has its output connected to the input of a divider circuit  5302 . The divider circuit  5302  generates a four bit code which is provided back to the RF oscillator via a four bit bus  5304 . The circuit described in  FIG. 53  has the advantage that it is synchronous. The rate of RF frequency change is locked to the RF carrier. However, the circuit includes a 2 GHz divider circuit that requires approximately 1 milliamp of Vdd current. 
         [0162]    The schematic diagram for this circuit is illustrated in  FIG. 54 . The schematic diagram of  FIG. 54  is similar to that described with respect to  FIG. 51  and like components are numbered in a similar fashion. The four bit codes are provided to the gates of a first group of transistors  5102  and a second group of transistors  5103  to turn the transistors on and off. Each of the four transistors in group  5102  has its source/drain path connected between a capacitor  5104  and ground. At the other end, each of the capacitors  5104  are connected to a node  5106 . Each of the transistors  5103  has its source/drain path connected between a capacitor  5108  and ground. The other side of each of the capacitors  5108  is connected to node  5110 . An additional capacitor  5112  is connected between node  5106  and ground. A capacitor  5114  is also connected between node  5110  and ground. 
         [0163]    Connected between nodes  5106  and  5110  is an inductor  5116 . A transistor  5118  is connected to the inductor  5116  at node  5110  and has its source/drain path connected between node  5110  and ground. The gate of transistor  5118  is connected to the opposite end of the inductor  5116  at node  5106 . Another transistor  5120  is connected to the inductor  5116  at node  5106 . The transistor  5120  has its source/drain path connected between node  5106  and ground. The gate of transistor  5120  is connected to the opposite end of inductor  5116  at node  5110 . Another transistor  5122  has its source/drain path connected between Vdd and node  5106 . The gate of transistor  5122  is connected to node  5120 . A final transistor  5124  has its source/drain path connected between Vdd and node  5110 . The gate of transistor  5124  is connected to node  5106 . An inverter  5126  is connected between node  5106  and the output node  5012  of the RF oscillator  5010 . Responsive to the control codes applied to the first and second groups of transistors  5102  and  5103 , the RF oscillator  5010  will generate a stepped RF carrier signal at its output  5012  between the first and second selected frequencies based upon values of the inductors and capacitors used within the circuit. This circuit additionally includes an inverter  5402  having its input connected to node  5110 . The output of the inverter  5402  is connected to a divider circuit  5302  which provides the four bit output to each of the transistor groupings  5102  and  5103 . 
         [0164]    Referring now to  FIG. 55 , there is illustrated a simulation of the resulting spectrum for an RF isolation link using a stepped frequency for the RF carrier signal as described herein above. As can be seen, there are generated sixteen separate peaks within the spectrum with an average power of approximately −24 dB for each peak. This illustrates the manner in which the emissions may be spread over sixteen separate frequencies rather than being concentrated on a single frequency when a single RF carrier signal is utilized. 
         [0165]    The circuits described in  FIGS. 50 and 53  for generating the frequency variation of the RF oscillator have the side effect of causing a tone within the emission spectrum if the isolator is used in an analog control loop such as switch controls in a switching power supply. Referring now to  FIG. 56 , there is illustrated an embodiment for using a random number generator to control generation of the code for providing the RF frequency. A 50-60 MHz ring oscillator  5602  provides an oscillation signal to a divide by 64 circuit  5604 . The output of the divider circuit  5604  is provided as the clock input to a 10-bit linear shift register  5606 . The linear shift register circuit  5606  may comprise the well known Debruijn counter circuit that prevents the register from becoming stuck in an all zero condition. The outputs of the 10-bit shift register  5606  are provided as input to a NOR gate  5608 . The b 0  bit and the b 9  bit output from the 10-bit shift register  5606  are provided as input to an exclusive OR gate  5610 . The output of the exclusive OR gate  5610  and the output of the NOR gate  5608  are provided as inputs to an exclusive OR gate  5612 . The output of the exclusive OR gate  5612  is provided as the data input to the 10-bit shift register  5606 . The RF oscillator circuit described with respect to  FIG. 51 , has its inputs connected to the b 0 , b 1 , b 2  and b 3  outputs of the 10-bit shift register  5606 . The RF oscillator circuit generates the stepped RF carrier signal in response to this 4-bit code input and generates an output RF carrier signal at output  5616 . 
         [0166]    Although the preferred embodiment has been described in detail, it should be understood that various changes, substitutions and alterations can be made therein without departing from the scope of the invention as defined by the appended claims.