PATENT ABSTRACT
There is disclosed a fractional-spaced equalizer (FSE) that is capable of performing joint intersymbol-interference (ISI) cancellation and matched filter (MF) processing. The FSE employs a constrained optimization technique to control the out-of-band frequency response of the equalizer&#39;s FIR while, at the same time, controlling the pass-band and roll-off of the FIR to cancel ISI. The format of the constrained optimization technique permits a single bank of multipliers elements to service the inner product computations associated both with the ISI cancellation and MF operations. This time-multiplexing technique promotes a conservation of hardware associated with the MF, and provides for a reduction in the computational complexity leading to an increase in power efficiency.

PATENT DESCRIPTION
TECHNICAL FIELD OF THE INVENTION  
         [0001]    This invention relates to digital equalizer technology and more particularly to fractional-spaced equalizers that are preceded by a MF.  
         BACKGROUND OF THE INVENTION  
         [0002]    In terms of suppressing out-of-band distortion and canceling multi-path ISI, the standard design of most digital receivers cascades a MF with an adaptive equalizer. The MF usually takes the form of a square-root raised cosine (RRC) response in order to maximize SNR while the adaptive equalizer often operates as a fractional-spaced equalizer (FSE) to allow inverse modeling of the propagation channel across the full spectral band, as apposed to only at pass-band frequencies in the symbol-spaced equalizer.  
           [0003]    In modern receivers the transmitter&#39;s pulse-shaping process re-proportions the spectral energy of the base-band signal such that the major spectral components occupy the low-frequency band while the minor spectral lobes are made to occupy the high frequency band. The band-limited representation of the transmitted signal infers that channel inversion at out-of-band frequencies is only as important to restoring full-received SNR as to the extent that the signal&#39;s high frequency spectral components are important to representing the distortion-less transmitted modulation.  
           [0004]    In terms of suppressing such out-of-band distortion signals as CW jamming the FSE has historically relied upon both the recursion of its ISI-canceling operation and on the out-of-band attenuation characteristics of the MF preceding it. If the source of the distortion is thermal noise, however, the FSE must rely exclusively on the MF as the equalizer&#39;s spectral side-lobe levels are not well defined, are susceptible to variations in the adaptation constant, and therefore, cannot suppress the noise prior to decimation of the signal to the symbol rate.  
           [0005]    With the importance of multi-path ISI cancellation relatively unimportant at out-of-band frequencies and with the pre-FSE RRC MF providing suppression of out-of-band interference, the responsibilities of the FSE&#39;s out-of-band mask have remained limited to the cancellation of excess adjacent channel interference not suppressed by the MF. It is noted that both the pre-FSE RRC MF and FSE are digital filters that operate at the same sample rate and whose responsibilities across the frequency band are approximately decoupled. Because of this, both filters can be combined into a single filter if control over the FSE&#39;s spectral mask can sustain well-defined side-lobes that are immune to changes in the equalizer&#39;s adaptation constant.  
           [0006]    This suggests a single filter implementation for the cascade design. From the point of hardware, a cascading of successive digital filters demands that separate bank of FPGA multipliers must be used to service the demands of each filter in the cascade chain. A single filter implementation of the traditional RRC MF plus FSE cascade design reduces cost as similar hardware components can be used to service the processing of associated with each. Although an increase in computational complexity must result when two separate processes are combined to conserve hardware, many options present themselves to minimize this increase. Consequently, reductions in both computational complexity and power consumption over that required for the traditional cascade design are possible.  
           [0007]    Accordingly, it is one objective of the present invention to provide a FSE that can simultaneously affect control over the equalizer&#39;s pass-band, roll-off, and side-lobe characteristics using a technique of constrained optimization so as to form a joint ISI-cancelling and MF update that can achieve the received SNR performance of the state-of-the-art cascade pre-FSE RRC MF plus FSE design.  
           [0008]    It is another object of the present invention to provide a FSE that implements a time-multiplexing architecture that enables a single bank of multiplier elements to perform the inner product computations associated with both the ISI-canceling and MF updates within the confines of a constrained optimization update, the purpose to provide for reduced hardware complexity over that of the state-of-the-art cascade pre-FSE RRC MF plus FSE design.  
           [0009]    It is the further object of the present invention to provide a FSE that operates as a joint ISI-canceling and MF FSE where the error associated with ISI-cancellation may be derived from any number of existing algorithms within the confines of the constrained optimization update of the present invention.  
           [0010]    It is the further object of the present invention to interject the process of time-domain windowing of the constraint waveform into the constrained optimization update so as to minimize the increase in computational complexity incurred from the introduction of the time-multiplexing architecture.  
           [0011]    It is the further object of the present invention to provide a FSE operating as a joint ISI-canceling and MF FSE where the rate at which the equalizer&#39;s weights are updated in accordance with the MF processing is controlled via an algorithm to minimize computational workload.  
           [0012]    It is the further object of the present invention to provide a FSE operating as a joint ISI-canceling and MF adaptive equalizer which implements an initialization of the equalizer&#39;s FF weights using a selected set of coefficients of an RRC MF, the intent of which is to reduce the acquisition time of the MF characteristics of the FSE&#39;s steady-state joint inverse channel and MF function.  
           [0013]    It is the further object of the present invention to provide a FSE operating as a joint and MF FSE under a constrained optimization update when the FSE is partitioned as a poly-phase process.  
         SUMMARY OF THE INVENTION  
         [0014]    These and other problems have been solved by combining the pre-FSE RRC MF and the FSE into a single filter joint process FSE that controls the spectral side-lobe behavior of the equalizer while simultaneously maintaining control over the equalizer&#39;s spectral pass-band and roll-off characteristics in accordance with the criterion for cancellation of ISI.  
           [0015]    To accomplish this, the gradient descent algorithm of the standard unconstrained FSE, which is responsible for driving the ISI-cancellation process, was modified to incorporate a constraint via the Lagrange multiplier technique. The constraint is defined to be a restriction that the equalizer&#39;s weights be orthogonal to a waveform whose major spectral components reside at out-of-band frequencies. If the FSE is operated as a base-band equalizer then out-of-band refers to the high frequency band, and if FSE is operated as a band-pass equalizer, out-of-band refers to that portion of the frequency band not occupied by the signal&#39;s major spectral components.  
           [0016]    To generate the orthogonality between the equalizer&#39;s weights and the out-of-band signal the inner product of these two time sequences is computed and subtracted from a desired orthogonality target on an iteration-by-iteration basis. The difference is scaled and then used to update the equalizer&#39;s weights in a direction so as to minimize the error. The orthogonality target is a scalar β of value less than 1.0 which forces the recursion of the constraint update to generate a low-pass process of the FSE at out-of-band frequencies. As a result, the spectral side-lobe development of the FSE is controlled so as to generate a low-pass process. The error between the measured orthogonality and targeted orthogonality is termed the constraint error and the high frequency signal is termed the constraint waveform.  
           [0017]    Thus, the modified FSE updates the equalizer&#39;s weights twice per equalizer iteration—once in accordance with the minimization of the mean-squared error associated with ISI cancellation and the second time in accordance with the criterion for the added constraint. The modified equalizer is termed the joint ISI-canceling and MF adaptive equalizer, and because of the development of well-defined spectral side-lobes at steady-state, this joint process equalizer provides for robust RRC MF processing as well as simultaneous channel inversion.  
           [0018]    The primary technical advantage of the present invention is that from the configuration of the linearly constrained algorithm driving joint process equalizer a time-multiplexing architecture is employed to allow a single bank of multiplier elements to service two inner product computations, that associated with the ISI cancellation and that associated with the added constraint for the MF processing. A single bank of multipliers servicing the needs of both ISI cancellation and MF processing contrasts with the two separate banks of multipliers required in the traditional cascade design, and therefore, the present invention conserves hardware in the system design.  
           [0019]    The present invention pertains to the creation of a joint ISI-canceling and MF process for any FSE that is sampled at more than 1-sample-per-symbol and satisfies the Nyquist criterion. Although a FSE that is sampled exactly at the Nyquist rate can sufficiently perform full band channel inversion, FSEs are often sampled above the Nyquist rate at 2-samples-per-symbol (or twice the transmitted symbol rate) in order to simplify the subsequent task of down-sampling to the symbol rate. Hence, a preferred embodiment of the present invention is designed to operate the constrained FSE at twice the symbol rate. In this case, the decimation device at the output of the equalizer takes the form of a 2:1 commutator.  
           [0020]    The time-multiplexing structure of the present invention remains unchanged regardless of the choice of the preferred embodiment for the FSE&#39;s sampling rate. For example, the constrained joint ISI-canceling and MF FSE operating at 2-samples-per-symbol performs a time-share of a single bank of multiplier elements with the samples of the two sequences accessing the multiplier bank, either the input data and the equalizer&#39;s weights or the equalizer&#39;s weights and the constraint waveform, separated in time at half-intervals of the transmitted symbol rate, or T sym /2. In an alternate embodiment, the joint ISI-canceling and MF FSE can be sampled at less than 2-samples-per-symbol if the Nyquist criterion is satisfied. Here, the time-multiplexing scheme is still based upon a time-share of a single bank of multiplier elements, but now with the samples of the two sequences accessing the multiplier bank separated in time at fractional intervals less than T sym /2. Thus, although the number of multiplier elements contained within the single bank increases as the FSE is sampled at larger rates, the time-multiplexing architecture of the present invention still provides for a conservation of hardware.  
           [0021]    As previously mentioned, the constrained update of the present invention works in conjunction with an update responsible for inversion of the propagation channel. Amongst the various existing configurations for the algorithm that generates the error associated with channel inversion the constrained update of present invention is generally used with, but is not exclusive to, three different channel inversion updates: a training sequence update, a decision-based update, and a statistically based update. Used in conjunction with the newly formed linearly constrained algorithm, these three updates, as well as others not mentioned here, form embodiments of the present invention.  
           [0022]    In the training sequence embodiment the error associated with ISI-cancellation is formed from the difference between the equalized signal and a known training sequence. In an alternate embodiment, a decision device, or slicer, forms the error as the difference between the equalized signal and the slicer&#39;s output, which is a quantized version of equalized signal. A third embodiment of the present invention forms this error as the scaled difference between the power of the equalized signal and a parameter describing the statistical properties of the original transmitted modulated signal.  
           [0023]    A decision-directed configuration for the channel inversion error algorithm is often used in conjunction with a decision-feedback update for robust cancellation of ISI. Hence, a preferred embodiment of the present invention uses the joint ISI-canceling and MF linearly constrained update within the configuration of a decision-feedback equalizer.  
           [0024]    Another technical advantage of the present invention is that the joint ISI-canceling and MF constrained update can be applied to a FSE that is partitioned as a poly-phase process. The polyphase decomposition of the standard FSE embeds the M:1 decimator normally occurring at the output of the equalizer to within the FSE using the Nobel Identity. As a result, the multiplier bank is partitioned into M multiplier sub-banks, each containing half the total multipliers of the non-partitioned multiplier bank. Although the workload per-output-point of the poly-phase partitioned FSE is greater than that of the non-poly-phase partitioned FSE by a factor of M, the poly-phase configuration commutates every M-th input sample of the distorted waveform to only one of the M sub-banks. Thus, the situation in the non-poly-phase FSE in which every input sample is processed by every equalizer weight is avoided in the poly-phase configured FSE. With regard to the present invention the linearly constrained poly-phase FSE is designed to form a joint ISI-canceling and MF process by computing M additional inner products associated with the constraint update. In the preferred embodiment of the present invention where M=2 the joint ISI-canceling and MF poly-phase FSE is able to conserve the hardware associated with the RRC MF&#39;s multiplier elements at the expense of four times the workload of the standard non-poly-phase configured unconstrained FSE.  
           [0025]    A further technical advantage of the present invention is that when the constraint waveform is defined as a frequency dependent sinusoid residing in the high frequency band, the linearly constrained update may be modified to determine the most minimal duty cycle of the sinusoid required to saturate performance. In this embodiment the time-series of the high frequency sinusoidal signal is time-domain windowed with a digital window function so as to render the samples farthest from the midpoint of the total duty cycle with negligible importance as compared to the most central samples. The net effect is that the samples of negligible amplitude need not contribute in the inner product computation associated with the constraint update. This reduces the computational complexity and improves power efficiency.  
           [0026]    It has been determined that the constraint, windowed or not windowed, need not contribute an update of the equalizer&#39;s weights at all iterations. Thus, the joint process equalizer may be designed with a switch to control at which iterations an update of the equalizer&#39;s weights associated with the MF processing is to occur. This also results in a reduction in the number of computations and the operational power needed to run the joint process equalizer.  
           [0027]    Since the equalizer&#39;s impulse response at steady-state is to be a composite of both the inverse channel model and RRC MF, initialization of the equalizer with the RRC MF taps attains half the final solution from the start. Hence, initialization reduces acquisition time of the MF characteristics, which allows the MF processing to be partially eliminated sooner in the update.  
           [0028]    The foregoing has outlined rather broadly the features and technical advantages of the present invention in order that the detailed description of the joint ISI-canceling and MF adaptive equalizer that follows may be better understood. Additional features and advantages of this joint process equalizer will be described hereinafter which form the subject of the claims of the invention. It should be appreciated by those skilled in the art that conception and the specific embodiment disclosed may be readily utilized as a basis for modifying or designing other structures for carrying out the same purpose of the present invention. It should also be realized by those skilled in the art that such equivalent constructions do not depart from the spirit and scope of the invention as set forth in the appended claims. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0029]    For a more complete understanding of the present invention, and the advantages thereof, reference is now made to the following descriptions taken in conjunction with the accompanying drawings, in which:  
         [0030]    [0030]FIG. 1 is a high-level block diagram of a state-of-the-art MF plus FSE cascade system;  
         [0031]    [0031]FIG. 2 is a composite high-level/detailed block diagram of a state-of-the-art MF plus FSE cascade (non-poly-phase partitioned) system;  
         [0032]    [0032]FIG. 3 is a detailed block diagram of a preferred embodiment of the algorithm responsible for adjusting the error associated with channel inversion of the state-of-the-art MF plus FSE cascade system;  
         [0033]    [0033]FIG. 4 is a detailed block diagram of a preferred embodiment of the weight update algorithm of the state-of-the-art MF plus FSE cascade system;  
         [0034]    [0034]FIG. 5 is a high-level/detailed block diagram of the joint ISI-canceling and MF adaptive equalizer operating at 2-samples-per-symbol (M=2) in accordance with a preferred embodiment of the present invention;  
         [0035]    [0035]FIG. 6 is a detailed block diagram of the algorithm that generates samples of the constraint waveform in accordance with a preferred embodiment of the present invention;  
         [0036]    [0036]FIG. 7 is a detailed block diagram of the constraint error adjustment algorithm in accordance with a preferred embodiment of the present invention;  
         [0037]    [0037]FIG. 8 is a detailed block diagram of the rate control algorithm for the constraint update of the equalizer weights in accordance with a preferred embodiment of the present invention;  
         [0038]    [0038]FIG. 9 is a detailed block diagram of the algorithm that generates the channel inversion error which utilizes a training sequence-based update in accordance with a preferred embodiment of the present invention;  
         [0039]    [0039]FIG. 10 is a detailed block diagram of the algorithm that generates channel inversion error which utilizes a decision-directed update in accordance with a preferred embodiment of the present invention;  
         [0040]    [0040]FIG. 11 is a detailed block diagram of the algorithm that generates the channel inversion error which utilizes a blind update via the constant modulus algorithm (CMA), in accordance with a preferred embodiment of the present invention;  
         [0041]    [0041]FIG. 12 is a detailed block diagram of the algorithm that calculates the modulus factor of the blind CMA update, in accordance with a preferred embodiment of the present invention;  
         [0042]    [0042]FIG. 13 is a detailed block diagram of the algorithm which performs a decision-feedback update of the joint ISI-canceling and MF equalizer in accordance with a preferred embodiment of the present invention;  
         [0043]    [0043]FIG. 14 is a detailed block diagram of the algorithm which performs a windowing of the constraint waveform using a single window function in accordance with a alternate embodiment of the present invention;  
         [0044]    [0044]FIG. 15 is a detailed block diagram of an of the algorithm which performs a windowing of the constraint waveform using a window derived from a composite of many window functions in accordance with an alternate embodiment of the present invention;  
         [0045]    [0045]FIG. 16 is a detailed block diagram of the algorithm that performs an initialization of the contents of the equalizer FF weight register bank using the pre-FSE RRC MF taps in accordance with a preferred embodiment of the present invention; and  
         [0046]    [0046]FIG. 17 is a detailed block diagram of the polyphase configuration of the joint ISI-canceling and MF adaptive equalizer in accordance with a preferred embodiment of the present invention. 
     
    
     DETAILED DESCRIPTION OF THE PRIOR ART  
       [0047]    Before discussing the joint ISI-canceling and MF adaptive digital equalizer, it will be useful to discuss the state-of-the-art equalizer and MF cascade design.  
         [0048]    There are no known joint ISI-canceling and MF adaptive digital equalizer configurations that permit control the spectral side-lobes of the FSE so as to generate a RRC MF state across the full spectral band of the equalizer, and so, the state-of-the-art is defined to be the design in FIG. 1, an RRC MF  101  cascaded with a FSE  102 .  
         [0049]    [0049]FIG. 2 depicts a detailed block diagram representation of FIG. 2 showing the state of the art MF plus FSE cascade filter system operating at M-samples-per-symbol (M times the symbol rate).  
         [0050]    The process is as follows. Samples of the distorted input waveform contained within a data register bank  201  associated with the RRC MF processing are shifted to the right by one register position. The sample of distorted waveform is input to the first (far left) position of the MF data register bank  201 . The contents of the MF data register bank  201  then engage the contents of an RRC MF weight register bank  202 , which stores the RRC MF coefficients, in an inner product computation using a bank of multiplier elements  203  and a bank of summation nodes  204 . This inner product IP represents a sample of the post-filtered pre-equalized signal.  
         [0051]    Next, the contents of the equalizer data register bank  205 , storing the previous samples of the post-filtered and pre-equalized signal, is shifted to the right by one register position. The sample of the previously computed inner product IP is input to the first register position of the equalizer data register bank  205 . The contents of the equalizer data register bank  205  then engage the contents of the equalizer weight register bank  206 , storing the current values of the equalizer&#39;s weights, in an inner product computation using a bank of multiplier elements  207  and a bank of summation nodes  208 . This inner product represents a sample of the equalized signal sampled at M-samples-per-symbol.  
         [0052]    The equalized sample is then passed to a M:1 commutator  209  which decimates the equalized signal to the symbol rate. The decimated signal is then passed to an algorithm ALG-ISI-ERR  210  that forms the error needed to drive the weights in accordance with criterion for ISI cancellation. Three different state-of-the-art configurations for ALG-ISI-ERR  210  are based upon a training sequence, a decision device, or a statistically based update. These configurations will be presented in detail during the description of the embodiments of the present invention.  
         [0053]    Next, the error from ALG-ISI-ERR is delivered to an algorithm ALG- 1   211  that transforms the error into an adjustment signal used to update the equalizer&#39;s weights in accordance with the criterion for channel inversion. ALG- 1   211  refers to one of any number of algorithms that can transform the error into a signal capable of controlling the adaptation of the equalizer&#39;s weights.  
         [0054]    One possible configuration for ALG- 1   211  is shown in FIG. 3 and consists of a complex conjugation operator  301 , a scalar adaptation constant μ  302 , and a multiplier  303 . The adaptation constant μ  302  multiplies the ISI equalization error to form a product representing an adjustment signal for the weight update associated with the channel inversion process. The operating range for the adaptation constant μ  302  is  
         0&lt;μ&lt;μ crit   [1] 
         [0055]    where the critical value μ crit  is inversely proportional to the energy of the samples of distorted signal contained with in the equalizer data register bank  201  (ref. FIG. 2) at the current iteration.  
         [0056]    Continuing with the process flow of FIG. 2, the adjustment signal produced at the output of ALG- 1   211  is then delivered to an algorithm ALG- 2   212  which updates the contents of the equalizer weight register bank  206 . ALG- 2   212  refers to one of any number of algorithms that can make use of an adjustment signal to update the contents of the equalizer weight register bank  206 .  
         [0057]    One possible configuration for ALG- 2   212  is shown in FIG. 4. The sample of adjustment signal formed from the output of ALG- 1   212  (ref. FIG. 2) multiplies the contents of the equalizer data register bank  205  (ref. FIG. 2) using a bank of multiplier elements  401 . The resulting bank of products is added to the current contents of the equalizer weight register bank  206  (ref. FIG. 2) using a bank of summing nodes  402 . The bank of sums produced is stored in the equalizer weight register bank  206  (ref. FIG. 2).  
         [0058]    As a specific example of the equalizer&#39;s FF weight update, the contents of the first (far left) register of the equalizer data register bank  205  (ref. FIG. 2) multiplies the adjustment signal produced from ALG- 1   212  (ref. FIG. 2) using the first (far left) multiplier of the multiplier bank  401 . The product adds to the first (far left) register position of the equalizer weight resister bank  206  (ref. FIG. 2) using first (far left) summing node of the bank of summing nodes  402  and the sum is stored in the first register position of the equalizer weight register bank  206  (ref. FIG. 2). The process continues with the updates associated with the next successive register positions of the equalizer weight register bank  206  (ref. FIG. 2).  
         [0059]    Retuning to the processing in FIG. 2, after updating the contents of the equalizer weight register bank, the entire processing of the MF and FSE cascade system repeats with the contents of the MF data register bank  201  shifted to the right by one position to prepare for the next pre-filtered sample to be input.  
       DETAILED DESCRIPTION OF THE INVENTION  
       [0060]    [0060]FIG. 5 depicts a detailed block diagram of the joint ISI-canceling and MF adaptive equalized in accordance with a preferred embodiment of the present invention. FIG. 5 indicates the time-multiplexing architecture in which a single multiplier bank  504  is utilized to perform two separate inner product computations, that associated with the channel inversion update and that related to MF processing. In this embodiment the equalizer operates at 2-samples-per-symbol (twice the symbol rate M=2);  
         [0061]    Elements currently residing within a data register bank  501  are shifted to the right by one position. A sample of the distorted input waveform is then loaded into the first position of the data register bank  501 . At time t_ 1 , after the current distorted sample enters the first data register position, switch S 1   502  is closed to allow the contents of the data register bank  501  to engage the contents of a second register bank  503 , storing values of the equalizer&#39;s feedforward (FF) weights, in an inner product using a bank of complex multiplier elements  504  and summation nodes  505 . This inner product represents the current sample estimate of the equalized signal. Samples of the equalized signal calculated from the previous inner product are then passed through a 2:1 commutator  506 , which discards every other sample to decimate to the symbol rate. The samples of decimated signal are then delivered to switch S 1   502 . Switch S 1   502  in this position prohibits the inner product samples associated with the MF processing from entering the signal processing chain associated with the channel inversion process. With switch S 1   502  closed, samples of the equalized signal are delivered to an algorithm ALG-ISI-ERR  507  to form the error signal associated with the channel inversion update.  
         [0062]    The error signal associated with the channel inversion is then delivered to an algorithm ALG- 1   508  that transforms the error into an adjustment signal used to update the equalizer&#39;s weights in accordance with the criterion for channel inversion. ALG- 1   508  refers to one of any number of algorithms that can control the adaptation of the equalizer&#39;s weights.  
         [0063]    [0063]FIG. 3 illustrates a detailed block diagram of ALG- 1   508  in accordance with a preferred embodiment of the present invention as previously described and discussed.  
         [0064]    Referencing FIG. 5, the adjustment signal formed at the output of ALG- 1   508  is then passed to switch S 1   502 . In this position, switch S 1   502  permits an update of the equalizer&#39;s weights when the adjustment signal is derived from the channel inversion process and prevents the channel inversion error adjustment signal from updating the weights when the present invention switches its mode of operation to the MF processing. With switch S 1   502  in a closed position, the adjustment signal derived from ALG- 1   508  is delivered to algorithm ALG- 2   509  which performs an update of the contents of the equalizer&#39;s FF weight register bank  503 . ALG- 2   509  refers to one of any number of algorithms that can control the adaptation of the equalizer&#39;s weights.  
         [0065]    [0065]FIG. 4 illustrates a detailed block diagram of ALG- 2   509  in accordance with a preferred embodiment of the present invention as previously described and discussed.  
         [0066]    This completes the ISI-cancellation process of the present invention for the current iteration.  
         [0067]    Next, the MF processing is initiated. A third register bank  510  of the present invention stores samples of a signal defining samples of the constraint waveform. This waveform represents a signal whose major spectral components reside in the high frequency band and pertains to any function that can become uncorrelated with the equalizer&#39;s impulse response within the constrained optimization algorithm so as to transform the equalizer&#39;s spectral side-lobes into a robust spectral mask.  
         [0068]    A preferred embodiment of the present invention defines the constraint waveform as a set of independent complex sinusoids, each residing above the quarter-sample rate. For example, suppose the i-th constraint sinusoid in the set {i=1,2, . . . , N} is defined as  
           c   i ( k )= A exp{U (2π f   i   k +φ i )}  [2] 
         [0069]    The terms A, f, and φ refer to the constraint sinusoid&#39;s amplitude frequency, and phase. The frequency changes with time (equalizer iteration n) through index i so as to sweep out the entire out-of-band frequency band.  
           i =( n− 1)( mod ) N+ 1  [ 3 ] 
         [0070]    The total number of independent sinusoids N is determined by trial and error tuning to maximize the full-received SNR. A preferred embodiment of the present invention spaces the complex tones at equidistant frequency intervals starting at the spectral nulls of the pre-FSE RRC MF and ending at the half-sampling rate.  
         [0071]    Other embodiments may increment the frequency as in a ramping function or FM sweep (linear variation with iteration index n), or higher order non-linear variations. The constraint waveform may also be defined to be a real sinusoid or a real cosinusoid. Sinusoidal amplitude A may take on any value above zero, A&gt;0, or may take on a time-varying format A(t) if desired. An arbitrary phase either random or deterministic φ can be included or can be derived from many phases φ i . With respect to the constraint vector sinusoidal tap index k ranges from 1 to L where L is the number of equalizer coefficients.  
         [0072]    Referring back to FIG. 5, after the contents of the equalizer FF weight register bank  503  have been updated using ALG- 2   509  in accordance with the criterion for channel inversion, samples of the signal representing the constraint waveform are loaded into the constraint register bank  510  using an algorithm ALG- 3   511 . ALG- 3   511  refers to one of any number of algorithms that can be used to define the contents of the constraint register bank  510 .  
         [0073]    [0073]FIG. 6 depicts a detailed block diagram of ALG- 3   511  in accordance with a preferred embodiment of the present invention. An index i is formed from an overflow counter  601  that cyclically counts between 1 and N, at the symbol rate, and indexes a register bank  602  storing the N out-of-band constraint frequencies. The overflow counter  601  is comprised of a scalar fixed to the value of  1   603 , a register  604  to store the current state of the increment, an overflow test  605  to reset the value of the counter register  604  to 1 after the count exceeds the value set by a parameter N  606  which the overflow test  605  uses to conduct its comparison, and a summing node  607  to perform the incrementation of the counter register  604 .  
         [0074]    At the conclusion of the ISI-cancellation update, overflow counter  601  increments index i to access the next frequency in the frequency array  602 . The selected frequency f i  is then passed to a sinusoidal generator  608  which generates samples of the complex sinusoid.  
         [0075]    Referencing FIG. 5, the samples of constraint waveform are then loaded into the constraint register bank  510 , but can also be passed first to an optional algorithm ALG- 4   512  when the constraint waveform is sinusoidal in nature to limit duty cycle as a means to reduce the computational complexity of the MF processing. The composition of ALG- 4   512  will be discussed later.  
         [0076]    Since the samples of each sinusoid are known apriori to the equalization, the constraint waveform may be loaded from ROM. Thus, an alternate embodiment of the ALG- 3   511  and ALG- 4   512  combination is a ROM lookup table.  
         [0077]    Continuing with the process flow in FIG. 5, at time t_ 2 , shortly after time t_ 1  and prior to the input of the next distorted sample to be processed, switch S 1   502  is opened and a second switch S 2   517  is closed to allow computation of a second inner product, this time between the contents of the constraint register bank  510  and the contents of the equalizer FF weight register bank  503 . Once again the bank of multiplier elements  504  and bank of summing nodes  505  are used in forming the inner product. This inner product represents a measurement of the orthogonality between the equalizer&#39;s weights and the complex constraint sinusoid residing at frequency f i  at the current iteration.  
         [0078]    The sample of the previous inner product between the equalizer&#39;s weights and the constraint waveform is then passed to the 2:1 commutator  506 , which discards every other sample. Switch S 2   517  is again encountered to deliver the orthogonality inner product to the subsequent algorithms involved in the MF processing. With switch S 2   517  closed the orthogonality inner product at the output of the 2:1 decimator  506  is subtracted from a parameter β  518 , termed the constraint level, using differencing node  519 . The constraint level β  518  defines the targeted strength of the orthogonality between the equalizer&#39;s weights and the constraint waveform. The operating range for β  518  is  
         0≦β≦1  [4] 
         [0079]    The difference, termed the constraint error, is passed to an algorithm ALG- 5   520  that controls the rate of acquisition of the constraint update. A preferred embodiment of the present invention sets β=0 to maximize the strength of the orthogonality built between the equalizer&#39;s weights and the constraint waveform. In turn, the differencing node  519  is not necessary and so is removed. ALG- 5   520  refers to any number of algorithms that can be used to dictate the rate of acquisition of the constraint update.  
         [0080]    [0080]FIG. 7 depicts a detailed block diagram of ALG- 5   520  in accordance with a preferred embodiment of the present invention. Samples of the MF constraint error are passed to a complex conjugation operator  701  and the result is scaled by a parameter α  702 , equal to the inverse of the number of samples of the complex constraint sinusoid waveform multiplied by the amplitude of the complex constraint sinusoid, using a multiplier element  703 . When the constraint sinusoid is of unit amplitude the multiplicative parameter α  703  reduces to 1/L. The scaled MF constraint error represents an adjustment signal for the equalizer weight update associated with the MF processing.  
         [0081]    In FIG. 5 the adjustment signal formed at the output of ALG- 5   520  is then passed to switch S 2   517 . In this position, switch S 2   517  permits an update of the contents of the equalizer&#39;s FF weight register bank  503  when the adjustment signal is derived from the MF constraint error and prevents the MF constraint error adjustment signal from updating the contents of the equalizer FF weight register bank  503  when the present invention switches its mode of operation to back to the channel inversion process.  
         [0082]    With switch S 2   517  closed the adjustment signal derived from the MF constraint error is passed to an optional algorithm ALG- 6   521  to control the rate at which the contents of the equalizer weight register bank  503  are to be updated in accordance the criterion for the MF constraint. ALG- 6   521  refers to any number of algorithms that can be used to dictate the update rate of the equalizer&#39;s weights in accordance with the criterion for the constraint.  
         [0083]    [0083]FIG. 8 depicts illustrates ALG- 6   521  in accordance with a preferred embodiment of the present invention. The adjustment signal for the MF constraint update is passed to a switch S 3   801  whose open/closed state is controlled by an overflow counter  802 . The overflow counter  802  is comprised of a scalar fixed to the value of  1   803 , a register  804  to store the current state of the increment, a summing node  805 , an overflow test  806 , switches S 4   807 , S 5   808  and S 6   809 , and three parameters, P 0   810 , P 1   811 , and P 1   812 , to dictate the maximum count of the overflow counter  802 .  
         [0084]    As the joint ISI-canceling and MF adaptive equalizer initiates processing the overflow counter  802  counts at the symbol rate from 1 to the value set by parameter P 0   810  whose value is accessed by the overflow counter  802  through switch S 4   807  which, at initial conditions, is in a closed position. Through the duration of this count, switch S 3   801  is in a closed position to allow the adjustment signal derived from the MF constraint error to update the contents of the equalizer FF weight register bank  503  (ref. FIG. 5). When the incremental count in the register  804  exceeds the value specified by P 0   810  the overflow test  806  results in a binary TRUE, and switches S 3   801  and S 4   807  are opened. Switch S 4   807  is then disabled from the processing and remains in an open state throughout the remainder of the joint ISI-canceling and MF update.  
         [0085]    The increment register  804  is then set to zero and switch S 5   808  is closed to allow the overflow test  806  to use the value given by parameter P 1   811  as the new threshold of the maximum count. As the update of the joint process equalizer continues, the overflow counter  802  starts counting again from 1, but counts this time to the value given by P 1   811 . During this counting switch S 3   801  remains in an open position to keep the constraint update from updating the equalizer weights.  
         [0086]    Switch S 3   801  remains in an open position until the value in the counter register  804  exceeds the value given by P 1   811  upon which the overflow test  806  results in a binary TRUE again which prompts switch S 3   801  to close to resume updates of the contents of the equalizer&#39;s FF weight register bank  503  (ref. FIG. 5) in accordance with the MF constraint criterion. At the same time that switch S 3   801  is closed, switch S 5   808  is opened and switch S 6   809  is closed to set the maximum count of the overflow counter  806  to the value given by P 2   812 .  
         [0087]    The increment register  804  is again set to zero and as the update of the joint process equalizer continues, the overflow counter  802  starts counting again from  1   803 , but counts this time to the value given by P 2   812 . During this counting switch S 3   801  remains in a closed position until the value in the counter register  804  exceeds the value given by P 2   812  upon which the overflow test  806  results in a binary TRUE again which prompts switch S 3   801  to open to halt the update equalizer&#39;s weights. At the same time that switch S 3   801  is opened, switch S 5   808  is closed and switch S 6   809  is opened to set the maximum count of the overflow counter  802  back to the value given by P 1   811 . This process continues with the increment register  804  is again set to zero switch S 3   801  controlling constraint update rate through switches S 5   808  and S 6   809 , and parameters P 1   811  and P 2   812 .  
         [0088]    Referencing FIG. 5 again, after passing through ALG- 6   521  the adjustment signal associated with the constraint error, and derived from ALG- 5   508 , is delivered to algorithm ALG- 2   509  which performs an update of the contents of the equalizer&#39;s FF weight register bank  503 , this time in accordance with the MF constraint criterion. ALG- 2   509  refers to one of any number of algorithms that can control the adaptation of the equalizer&#39;s weights.  
         [0089]    [0089]FIG. 4 illustrates a detailed block diagram of ALG- 2   509  in accordance with a preferred embodiment of the present invention as previously described and discussed.  
         [0090]    This completes the MF portion of the joint ISI-canceling and MF operation at the current iteration.  
         [0091]    The entire joint process equalizer update is repeated for the next iteration beginning with the elements currently residing within the data register bank  501  being shifted to the right by one position and the next sample of the distorted input waveform being loaded into the first position of the data register bank  501 . The ISI-cancellation process is initialized once again with closure of switch S 1   502  and the computation the inner product between the contents of the data register bank  501  and the contents equalizer FF weight register bank  503 .  
         [0092]    We now discuss several possible configurations for ALG-ISI-ERR  507  of FIG. 5 which generates the error sequence that drives the update of the equalizer&#39;s weights in accordance with the criterion for the cancellation of ISI. ALG-ISI-ERR  507  pertains to any state-of-the-art algorithm that can derive an error signal pertaining to the cancellation of ISI.  
         [0093]    [0093]FIG. 9 depicts a detailed block diagram of ALG-ISI-ERR  507  of FIG. 5 in accordance with a preferred embodiment of the present invention. This embodiment utilizes a training sequence  901  to form the ISI error signal.  
         [0094]    With respect to FIG. 9, after passing through the 2:1 commutator  506  (ref. FIG. 5) and switch S 1   502  (ref. FIG. 5) the equalized signal is subtracted from a known training sequence  901 , which represents samples of the distortion-less transmitted sequence at symbol rate, using a differencing node  902 . The difference signal represents the ISI error sequence and is passed to ALG- 1   508  (ref. FIG. 5) to generate the adjustment signal that directs the equalizer&#39;s weight in accordance with ISI cancellation.  
         [0095]    [0095]FIG. 10 depicts a detailed block diagram of ALG-ISI-ERR  507  of FIG. 5 in accordance with an alternate embodiment of the present invention.  
         [0096]    With respect to FIG. 10, after passing through the 2:1 commutator  506  (ref. FIG. 5) and switch S 1   502  (ref. FIG. 5) the equalized signal is passed through a slicer (decision device)  1001  which quantizes the equalized signal to the closest 2-tuple of a decision region. A differencing node  1002  subtracts the pre-quantized sample from the quantized sample to form the error sequence which is then passed onto ALG- 1   508  (FIG. 5) to generate the equalizer weight adjustment signal. This embodiment is termed the decision-directed embodiment.  
         [0097]    [0097]FIG. 11 depicts a detailed block diagram of ALG-ISI-ERR  507  of FIG. 5 in accordance with an alternate embodiment of the present invention. In this embodiment the channel inversion error is formed via the use of a statistical-based update.  
         [0098]    With respect to FIG. 11, an algorithm ALG- 7   1101  computes the value of a parameter R m  measuring a ratio of statistical moments of the original modulated signal. A switch S 7   1102  is closed to deliver the value of parameter R m  to a register  1103  where it will reside throughout the processing. Switch S 7   1102  is then opened. Therefore, ALG- 7   1101  executes only a single time to calculate R m  and then is removed from the processing when switch S 7   1102  is opened.  
         [0099]    After passing through the 2:1 commutator  506  (ref. FIG. 5) and switch S 1   502  (ref. FIG. 5) the input sample of equalized signal is delivered to a complex conjugation device  1104 . The output of this complex conjugation device is then multiplied by the input sample of equalized signal using a multiplier device  1105 . The content of the register  1103  storing the value of parameter R m  is subtracted from this product via a differencing node  1106  and this difference is then multiplied by the input sample of equalized signal via a multiplier element  1107 . This last product represents a sample of the error sequence delivered to ALG- 1   508  (ref. FIG. 5) to form the weight adjustment signal as in previous embodiments.  
         [0100]    The statistically based parameter R m  is the ratio of moments of the amplitudes a j {j=1, 2, . . . , B} of the B-ary pre-pulse-shaped modulated constellation.  
           R   m   =E[|a   j | 2m   ]/E[|a   j | m ]  [3] 
         [0101]    ALG- 7   1101  refers to one of any number of algorithms that can compute the ratio of E[|a j | 2m ] to E[|a j | m ] where the operator E [x] denotes the expectation of x.  
         [0102]    [0102]FIG. 12 depicts a detailed block diagram of ALG- 7   1101  of FIG. 11 in accordance with a preferred embodiment of the present invention. The first amplitude a 1  of the set of amplitudes a j {j=1, 2, . . . , B} of the B-ary pre-pulse-shaped modulated constellation is passed through an absolute value operator  1201  to produce a value V 1 . A counter  1202  increments the value in a register  1203  from 0 to 1 with the use of a scalar  1204  set to the value of 1 and a summing node  1205 . A scalar  1206  set to the value of B and an overflow test  1207  test whether the contents in register  1203  have exceeded the value given by parameter B  1206 .  
         [0103]    V 1  is passed though a power operator  1208  which computes V 1  to the m-th power with use of parameter m  1209 . The output V 2  is then sent to two different paths of processing, an upper and lower path. In the upper path V 2  is passed through a squaring operation  1210  to form V 3  which, in turn, is delivered to the combination of a summing node  1211  and a delay register  1212  to perform an accumulation of future V 3  values. In the lower path V 2  is delivered to the combination of a summing node  1213  and delay register  1214  to perform an accumulation of future V 2  values. The entire process is repeated with input of the second 2-tuple a 2 .  
         [0104]    After all of the constellation 2-tuples have been processed the value in the increment register  1203  increments one more time. At this point, the value in the increment register  1203  exceeds the value given by parameter B  1206  and a CLOSE signal is sent to switches S 8   1215  and S 9   1216 . A division operator  1217  forms the ratio of the final values of V 2  and V 3  to form R m .  
         [0105]    [0105]FIG. 13 depicts a detailed block diagram of an alternate embodiment of the present invention. This embodiment closely resembles decision-directed embodiment with the exception that a feedback signal formed from a weighted set of previous slicer decisions adds to the equalized signal. This embodiment of the present invention is termed the decision-feedback embodiment.  
         [0106]    The process for the decision-feedback embodiment is as follows. With the output of the 2:1 commutator  506  (ref. FIG. 5) already formed and passed through switch S 1   502  (ref. FIG. 5), the contents of a decision register bank  1301 , which stores a set of the previous decisions produced from a slicer  1302 , engage the contents of a weight register bank  1303 , storing values of set of DF weights, in an inner product computation using a bank of multiplier elements  1304  and summing nodes  1305 . This inner product computation represents a sample of the DF&#39;s contribution to the total equalized signal.  
         [0107]    Next, the incoming signal to the DF embodiment is added to the DF sample using a summation node  1306 . The result of the addition represents a sample of the equalized signal and is passed to the slicer device  1302 . The equalized sample is subtracted from decision produced from the output of the slicer  1302  via a differencing node  1307  forming a sample of the ISI-cancellation error sequence. The error sample is then delivered to ALG- 1   508  (ref. FIG. 5) to form the adjustment signal needed to update the contents of the equalizer&#39;s FF weight register bank  501  (ref. FIG. 5).  
         [0108]    The sample of adjustment signal formed from ALG- 1   508  then multiplies contents of the decision register bank  1301  using a bank of multiplier elements  1308 . The resulting bank of products then adds to the current contents of the DF weight register bank  1303  using a bank of summing nodes  1309 , and the result is stored in the DF weight register bank  1303 . As an example of the DF weight update, the contents of the first (far left) register of the decision register bank  1301 , storing a set of previous decisions, multiplies the adjustment signal produced from ALG- 1   508 . The product adds to the first (far left) register position of the DF weight resister bank  1303  and the sum is stored in the first register position of the DF weight register bank  1303 . The update process continues with the next successive register positions of the DF weight register bank  1303 .  
         [0109]    At the completion of DF weight update the contents of the decision register bank  1301  are shifted to the right by one register position. The slicer  1302  output is delivered to first register position of the decision register bank  1301 . The DF operation then repeats with the inner product of the decision register bank  1301  and the DF weight register bank  1303 .  
         [0110]    With the description of the basic processing of the joint ISI-canceling and MF adaptive equalizer completed we now return to descriptions of both ALG- 4   512  and ALG- 7   522 .  
         [0111]    ALG- 4   512  performs windowing of the constraint waveform produced from ALG- 3   511  at the current equalizer iteration. When the constraint waveform is sinusoidal in nature a windowing of the sinusoidal time series weights the non-causal and causal samples furthest from the midpoint of the total duty cycle with negligible amplitude (or zero amplitude depending upon the selected window function) while emphasizing those samples nearest the duty cycle midpoint with greater importance. As a result, the samples of the windowed constraint waveform of negligible amplitude offer negligible contribution to the inner product between the equalizer FF weights and constraint waveform associated with the MF processing. Hence, the their multiply operations need not be performed and the inner product reduces to performing only the central-most multiply operations that will restore a measure of performance equivalent to that of the non-windowed constraint waveform.  
         [0112]    [0112]FIG. 14 illustrates a detailed block diagram of ALG- 4   512  in accordance with a preferred embodiment of the present invention. This embodiment performs windowing of the constraint waveform using a single window function. Samples of the constraint waveform produced from ALG- 3   511  (ref. FIG. 5) are passed to a multiplier bank  1401  which performs a point-by-point multiplication with contents of a register bank  1402  representing the window function. The resulting samples of windowed time series are then loaded into the constraint register bank  510  (ref. FIG. 5) and as the MF processing is initiated. However, in the inner product computation between the contents of the equalizer FF weight register bank  503  (ref. FIG. 5) and the contents of the constraint register bank  510  (ref. FIG. 5), only those central multiplications of the multiplier bank  504  (ref. FIG. 5) and central sums of the bank of summing nodes  505  (ref. FIG. 5) which correspond to the central samples of the windowed constraint waveform samples of non-zero or appreciable amplitudes are performed.  
         [0113]    For the singular windowing waveform various types of window functions may suffice for truncating the constraint waveform time series. For example, an unweighted window contains a steep decay in its time series which maximizes the number of extreme causal and non-causal samples of the windowed waveform that are of zero or negligible amplitude. This, in turn, minimizes the number of central-most multiplications and sums that need be performed in the inner product associated with the MF constraint. The drawback, however, is that windowing with an unweighted function maximizes the amount of spectral leakage induced from waveform time series truncation which diminishes the strength of the orthogonality between equalizer and constraint waveform.  
         [0114]    To compensate, the window function selected may contain a gradual decay of its time response such as the Hann, Hamming, Kaiser, etc. windows. For these windows, however, the number of samples of the windowed waveform, which are of zero or negligible amplitude, may not result in an appreciable workload reduction of the MF constraint processing. Hence, the window waveform is formed from a composite of multiple windows to achieved desired time series truncation with minimal spectral leakage.  
         [0115]    [0115]FIG. 15 illustrates a detailed block diagram of ALG- 4   512  in accordance with an alternate embodiment of the present invention. This embodiment performs windowing of the constraint waveform using multiple window functions. Switch S 8   1501 , initially in a closed position, allows the entire set of samples of the constraint waveform produced from ALG- 3   511  (ref. FIG. 5) to be loaded to a register bank  1502 . Switch S 8   1501  is then opened. Next, a bank of multiplier elements  1503  performs a point-by-point multiplication of the pre-windowed samples contained in register bank  1502  with samples of the first windowing function contained in a register bank  1504 . The bank of products is stored in register bank  1502 . Switch S 11   1508  remains in an open position until all windowing waveforms have been utilized. Switch S 12   1505  then moves to the windowing waveform #2 register bank  1506 . The bank of multiplier elements  1503  performs a point-by-point multiplication of the pre-windowed samples contained in register bank  1502  with samples of the second windowing function contained in register bank  1506 . The bank of products is stored in register bank  1502 . The process is repeated for all the windowing functions up to an including the last storing in windowing waveform #W register bank  1507 . Then switch S 11   1508  is closed to send the final version of the samples of windowed constraint waveform to the constraint register bank  510  (ref. FIG. 5).  
         [0116]    ALG- 7   522  is now discussed. Referencing FIG. 5, ALG- 7   522  performs an initialization of the contents of the equalizer FF weight register bank  503  with the coefficients of the RRC MF as a means to decrease the acquisition time needed to form the MF characteristics of the equalizer&#39;s composite inverse channel and MF function.  
         [0117]    [0117]FIG. 16 illustrates a detailed block diagram of ALG- 5   522  in accordance with a preferred embodiment of the present invention. Prior to the contents of the data register bank being shifted to prepare for the first sample of the distorted waveform to be loaded into the first position of the data register bank  501  (ref. FIG. 5), switch S 13  is closed to allow a coefficient set  1601  of the pre-FSE RRC MF, which spans the duration of the equalizer&#39;s FF weights, to be loaded into the register positions of the equalizer FF weight register bank  503  (ref. FIG. 5). Switch S 12   1602  is then opened to disconnect ALG- 7   522  (ref. FIG. 5) from the joint process equalizer update.  
         [0118]    [0118]FIG. 17 depicts a detailed block diagram of the joint ISI-cancelling and MF adaptive equalized partitioned as a polyphase process in accordance with a preferred embodiment of the present invention. The 2:1 decimator (ref. FIG. 5), previously at the output of the equalizer, is embedded within the fractional-spaced equalizer via the Nobel Identity. As a result, the data register bank  502  (ref. FIG. 5) is partitioned into two sub-register banks, u 0    1701  and u 1    1702 , each of which is half the length of the original non-polyphase partitioned data register bank  502  (ref. FIG. 5). In a similar manner the equalizer weight register bank  503  (ref. FIG. 5) is partitioned into sub-register banks w 0    1703  and w 1    1704 , and the constraint register bank  510  (ref. FIG. 5) is partitioned into sub-register banks c 0    1705  and c 1    1706 .  
         [0119]    The polyphase process of FIG. 17 is as follows. The elements contained within the data sub-register bank u 1    1702  are shifted to the right by one position. A sample of the distorted input waveform is then delivered to the first register position of data sub-register bank u 1    1702  via a 2:1 commutation device COM_ 1   1707 .  
         [0120]    At time t_ 1 , a switch S 1   1708  is closed to allow the contents of the data sub-register bank u 1    1702  to engage the contents of equalizer FF weight sub-register bank w 1    1704 , storing half of the equalizer&#39;s weights, the odd indexed weights (or even indexed weights depending upon polyphase methodology), in an inner product computation using a bank of multiplier elements  1709  and a bank of summation nodes  1710 .  
         [0121]    In this embodiment, the polyphase configuration of the present invention, the number of multiplier elements in the multiplier bank  1709  and number of summing nodes in the bank of summing nodes  1710  are both approximately half that of each of the non-polyphase configuration. The inner product computation represents half the total equalized decimated result at the current iteration, and with switch S 1   1708  closed the inner product is stored in a single delay element  1711  for future use. Switch S 14   1712 , currently in an open position, prevents the error sample associated with ISI cancellation from being formed until the second half of the total equalized decimated signal is computed.  
         [0122]    Next, samples of the signal representing the constraint waveform are loaded into constraint sub-register banks c 0    1705  and c 1    1706  using algorithm ALG- 3   1728 , but as in the non-polyphase embodiment, can also be passed first to an optional algorithm ALG- 4   1729  when the constraint waveform is sinusoidal in nature to limit duty cycle as a means to reduce the computational complexity of the MF processing. FIG. 6 illustrates a detailed block diagram of ALG- 3   1728  in accordance with a preferred embodiment of the present invention as previously described and discussed. FIGS.  14 - 15  illustrate detailed block diagrams of ALG- 4   1729  in accordance with preferred embodiments of the present invention as previously described and discussed.  
         [0123]    At time t_ 2 , shortly after time t_ 1  and prior to the input of the next distorted sample to be processed, switch S 1   1708  is opened and a second switch S 2   1713  is closed to allow computation of a second inner product, this time between the contents of the constraint sub-register bank c 1    1706  and the contents of the equalizer FF weight sub-register bank w 1    1704 . Again the bank of multiplier elements  1709  and bank of summing nodes  1710  are used in forming this inner product. This second inner product represents the first half the total contribution to the sample measuring the orthogonality between the equalizer&#39;s FF weights and the constraint waveform at the current iteration. With switch S 2   1713  closed it is stored in a single delay element  1714  for future use. Switch S 15   1715 , currently in an open position, prevents the MF constraint error from being formed until the second half of the total contribution to the measure of orthogonality between the equalizer&#39;s FF weights and the constraint waveform is formed. Switch S 2   1713  is then opened.  
         [0124]    Next, commutator COM  1   1707  moves to the data sub-register bank u 0    1702 , and a new sample of the distorted waveform is input to the first register position of data sub-register bank u 1    1702 . A second commutator COM  2   1716  moves to data sub-register bank u 0    1701  and constraint sub-register bank c 0    1705 , while a third commutator COM  3   1717  moves to the equalizer FF sub-register bank w 0    1703 , with all commutator movements controlled by a clock  1718 .  
         [0125]    The elements contained within the data sub-register bank u 0    1701  are then shifted to the right by one position and the next sample of the distorted input waveform is then delivered to the first register position of data sub-register bank u 0    1701  via the commutation device COM_ 1   1707 .  
         [0126]    At time t_ 3 , shortly after the input of the next distorted sample to data sub-register u 0    1701 , switch S 1   1708  is closed to allow the contents of the data sub-register bank u 0    1701  to engage the contents of equalizer FF weight sub-register bank w 0    1703  in a third inner product computation using the bank of multiplier elements  1709  and bank of summing nodes  1710 . With switch S 1   1708  closed this third inner product is added to the first inner product currently stored in the delay register  1711  via a summing node  1719  to form a sum Ps 1 . Ps 1  represents a sample of the equalized signal at the current iteration. Next, switch S 14   1712 , is closed to send sum Ps 1  to ALG-ISI-ERR  1720  to form the error associated with ISI-cancellation. FIGS.  9 - 11  illustrate detailed block diagrams of ALG-ISI-ERR  1720  in accordance with preferred embodiments of the present invention as previously described and discussed.  
         [0127]    The error is then sent to ALG- 1   1721  to form the adjustment signal needed to update the equalizer&#39;s FF weights in accordance with the criterion for ISI cancellation. FIG. 3 illustrates a detailed block diagram of ALG- 1   1721  in accordance with a preferred embodiment of the present invention as previously described and discussed.  
         [0128]    Continuing with the process description of FIG. 17 the adjustment signal formed at the output of ALG- 1   1721  is then passed to switch S 1   1708 . In this position, switch S 1   1708  permits an update of the contents of the equalizer&#39;s FF weights when the adjustment signal is derived from the channel inversion error and prevents the channel inversion error adjustment signal from updating the FF weights when the present invention switches its mode of operation to the MF processing. With switch S 1   1708  in a closed position, the adjustment signal derived from ALG- 1   1721  is delivered to algorithm ALG- 2   1722  which performs an update of the contents of the equalizer FF weight sub-register banks w 0    1703  and an update of the contents in equalizer FF weight sub-register banks w 1    1704 . FIG. 4 illustrates a detailed block diagram of ALG- 2   1722  in accordance with a preferred embodiment of the present invention as previously described and discussed.  
         [0129]    This completes the weight update associated with the ISI-cancellation process at the current iteration for polyphase embodiment of the present invention.  
         [0130]    At time t_ 4 , shortly after time t_ 3  and prior to the input of the next distorted sample to be processed, switch S 1   1708  is opened and switch S 2   1713  is closed to allow computation of an inner product, the fourth inner product in the series, between the contents of the constraint sub-register bank c 0    1705  and the contents of the equalizer FF weight sub-register bank w 0    1703  using the bank of multiplier elements  1709  and the bank of summing nodes  1710  servicing the computation. With switch S 2   1713  closed the fourth inner product is added to the second inner product currently stored in the delay register  1714  via a summing node  1723  to form a sum Ps 2 . Ps 2  represents a sample of the total measure of orthogonality between the equalizer&#39;s FF weights and constraint waveform at the current iteration.  
         [0131]    Next, switch S 15   1715  is closed to send sum Ps 2  onto the processing that derives the constraint error. Ps 2  is subtracted from the constraint level parameter β  1724  using differencing node  1725  as in the previous embodiments. The difference, termed the constraint error, is passed to an algorithm ALG- 5   1726 , as in prior embodiments, to form the adjustment signal needed to update the contents of both equalizer FF weight sub-register banks, w 0    1703  and W 1    1704 , in accordance with the MF constraint criterion.  
         [0132]    With switch S 2   1713  closed the adjustment signal derived from the MF constraint error is passed to algorithm optional ALG- 6   1727  to control the rate at which the contents of equalizer FF weight sub-register banks w 0    1703  and w 1    1704  are to be updated in accordance with the MF constraint criterion. FIG. 8 illustrates a detailed block diagram of ALG- 6   1722  in accordance with a preferred embodiment of the present invention as previously described and discussed.  
         [0133]    Continuing with the process description of FIG. 17, after passing through ALG- 6   1727  the adjustment signal associated with the constraint error, and derived from ALG- 5   1726 , is delivered to algorithm ALG- 2   1722 , as in previous embodiments, which performs an update of the contents of the both the equalizer FF weight sub-register banks w 0    1703  and w 1    1704  in accordance with the MF constraint criterion. Switch S 15   1715  is then opened.  
         [0134]    This completes the MF portion of the joint ISI-cancelling and MF operation at the current iteration for the polyphase embodiment of the present invention.  
         [0135]    Commutator COM_ 1   1707  then moves back to the u 1  register bank  1702 , COM_ 2   1716  moves back to data sub-register bank u 1    1702  and constraint sub-register bank c 1    1706 , and COM_ 3   1717  moves back to the equalizer FF weight sub-register bank w 1    1704 , all movements again controlled by the clock  1718 . Also the contents of delay register  1711  and delay register  1714  are zeroed out.  
         [0136]    The entire process is repeated with the contents equalizer data sub-register bank u 1    1702  shifted to the right by one position to prepare for the next sample of distorted input waveform to be delivered to the first register position of u 1    1702 .  
         [0137]    With respect to the configuration of ALG-ISI-ERR  1720  in the polyphase embodiment, FIGS.  9 - 11  illustrate detailed block diagrams of ALG-ISI-ERR  1720  in accordance with preferred embodiments of the present invention as previously described and discussed.  
         [0138]    When ALG-ISI-ERR  1720  in the polyphase embodiment is defined as in FIG. 10, where a slicer forms the equalization error, a DF configuration can be used to enhance the cancellation of ISI. FIG. 12 illustrates a detailed block diagram of the DF configuration in accordance with a preferred embodiment of the present invention as previously described and discussed.  
         [0139]    The polyphase embodiment of the present invention also benefits from initialization of the equalizer&#39;s FF weights using ALG- 7   1730 . FIG. 15 illustrates a detailed block diagram of ALG- 7   1730  in accordance with a preferred embodiment of the present invention. Since the constraint register bank  510  (ref. FIG. 15) is partitioned into sub-register banks c 0    1705  and c 1    1706  in the polyphase embodiment, ALG- 7   1730  initializes sub-register banks c 0    1705  and c 1    1706  with the odd and even indexed RRC MF taps, respectively, or vice versa.  
         [0140]    Although the present invention and its advantages have been described in detail, it should be understood that various changes, substitutions and alterations can be made herein without departing from the spirit and scope of the invention as defined by the appended claims.