Patent Application: US-201013581629-A

Abstract:
disclosed are methods for selecting auditory signal components for reproduction by means of one or more supplementary sound reproducing transducers , such as loudspeakers , placed between a pair of primary sound reproducing transducers , such as left and right loudspeakers in a stereophonic loudspeaker setup or adjacent loudspeakers in a surround sound loudspeaker setup . also disclosed are devices for carrying out the above methods and systems of such devices .

Description:
in the following , a specific embodiment of a device according to the invention , also termed a stereo to multi - mono converter , is described . in connection with the detailed description of this embodiment , specific numerical values for instance relating to respective angles in the loudspeaker set - up are used both in the text , figures and occasionally in various mathematical expressions , but it is understood that such specific values are only to be understood as constituting an example and that other parameter values will also be covered by the invention . the basic functional principle of this converter will be described with reference to the schematic block diagram shown in fig9 . while the embodiment shown in fig9 is scalable to n loudspeakers , and can be applied to auditory scenes encoded with more than two channels , the embodiment described in the following provides extraction of a signal for one supplementary loudspeaker in addition to the left and right loudspeakers ( the “ primary ” loudspeakers ) of the normal stereophonic reproduction system . as shown in fig1 , the one supplementary loudspeaker 56 is in the following detailed description generally placed rotated relative to the 0 ° azimuth direction and in the median plane of the listener . the scenario shown in fig1 constitutes one specific example , wherein v listen is equal to zero degrees azimuth . referring again to fig9 , the stereo to multi - mono converter ( and the corresponding method ) according to this embodiment of the invention comprises five main functions , labelled a to e in the block diagram . in function block a , a calculation and analysis of binaural signals is performed in order to determine if a specific signal component in the incoming stereophonic signal l source [ n ] and r source [ n ] ( reference numerals 14 and 15 , respectively ) is attributable to a given azimuth interval comprising the supplementary loudspeakers 56 used to reproduce the audio signal . such an interval is illustrated in fig1 and 11 corresponding to the centre loudspeaker 56 . the input signal 14 , 15 is in this embodiment converted to a corresponding binaural signal in the hrtf stereo source block 24 and based on this binaural signal , interaural level difference ( ild ) and interaural time difference ( itd ) for each signal component in the stereophonic input signal 14 , 15 are determined in the blocks termed ild music 29 and itd music 30 . in boxes 25 and 26 , the left and right angle limits , respectively , are set ( for instance as shown in fig1 and 11 ) based on corresponding input signals at terminals 54 ( left range ), 53 ( listening direction ) and 55 ( right range ), respectively . the corresponding values of the hrtf &# 39 ; s are determined in 27 and 28 . these hrtf limits are converted to corresponding limits for interaural level difference and interaural time difference in blocks 31 , 32 , 33 and 34 . the output from functional block a ( reference numeral 19 ) is the ild and itd 29 , 30 for each signal component of the stereophonic signal 14 , 15 and the right and left ild and itd limits 31 , 32 , 33 , 34 . these output signals from functional block a are provided to the mapping function in functional block c ( reference numeral 21 ), as described in the following . the input stereophonic signal 14 , 15 is furthermore provided to a functional block b ( reference numeral 20 ) that calculates the inter - channel coherence between the left 14 and right 15 signals of the input stereophonic signal 14 , 15 . the resulting coherence is provided to the mapping function in block c . the function block c ( 21 ) maps the interaural differences and coherence calculated in the function a ( 19 ) and b ( 20 ) into a filter d ( 22 ), which interaural differences and inter - channel coherence will be used to extract those components of the input signals l source [ n ] and r source [ n ] ( 14 , 15 ) that will be reproduced by the centre loudspeaker . thus , the basic concept of the extraction is that stereophonic signal components which with a high degree of probability will result in a phantom source being perceived at or in the vicinity of the position , at which the supplementary loudspeaker 56 is located , will be routed to the supplementary loudspeaker 56 . what is meant by “ vicinity ” is in fact determined by the angle limits defined in block a ( 19 ), and the likelihood of formation of a phantom source is determined by the left and right inter - channel coherence determined in block 20 . the basic functions of the embodiment of the invention shown in fig9 are described in more detail below . the specific calculations and plots relate to an example wherein a signal is extracted for one additional loudspeaker placed at zero degrees azimuth between a left and right loudspeaker placed at +/− 30 degrees azimuth , respectively , this set - up corresponding to a traditional stereophonic loudspeaker set - up as shown schematically in fig1 . the corresponding values of the left range , listening position , and right range input signals 54 , 53 , 55 are here chosen to be − 10 degrees , 0 degrees , + 10 degrees azimuth , corresponding to the situation shown in fig1 . the first step consists of calculating ear input signals l ear [ n ] and r ear [ n ] by convolving the input stereophonic signals l source [ n ] and r source [ n ] from the stereo signal source with free - field binaural impulse responses for sources at − 30 ° ( h − 30 ° l [ n ] and h − 30 ° r [ n ]) and at + 30 ° ( h + 30 ° r [ n ] and h + 30 ° l [ n ]). time - domain convolution is typically formulated as a sum of the product of each sample of the first sequence with a time reversed version of the other second sequence shown in the following expression : these signals correspond to the ear input signals in the case of ideal stereophony as described above . the centre loudspeaker is intended to reproduce a portion of the auditory scene that is located between the left angle limit , v llimit , and the right angle limit , v rlimit that are calculated from the angle variables left range , right range and listening direction ( also referred to as v lrange , v rrange and v listen ) as in the following equations : in the present specific example , v lrange , v rrange are −/+ 10 degrees , respectively , and v listen is 0 degrees . if the playback system contains multiple loudspeakers , then the angle variables left range , right range and listening direction allow the orientation and width of the rendered auditory scene to be manipulated . fig1 shows an example where listening direction is not zero degrees azimuth with the result being a rotation of the auditory scene to the left when compared to the scenario in fig1 . changes to these variables could be made explicitly by a listener or could be the result of a listener position tracking vector ( for instance a head - tracker worn by a listener ). furthermore , in fig3 there is shown a more general situation , in which the listening direction is outside the angular range comprising the supplementary loudspeaker 56 . although not described in detail , this situation is also covered by the present invention . the ild and itd limits in each case are calculated from the free - field binaural impulse responses for a source at v llimit degrees , h vllimitdegl [ n ] and h vllimitdegr [ n ], and a source at v rlimit degrees , h vrlimitdegl [ n ] and h rlimitdegr [ n ]. in the present embodiment , the remainder of the signal analysis in functions a through d operates on frequency domain representations of blocks of n samples of the signals described above . a rectangular window is used . in the examples described below n = 512 . the frequency domain representations of a block of the ear input signals , music signals and the binaural impulse responses ( for a source in the free - field at 0 °— this processing is for the centre loudspeaker ) are calculated using the dft as formulated in the equations below : as mentioned above , ild leftlimit , ild rightlimit and ild music are calculated from the magnitude of the appropriate transfer function . similarly , itd leftlimit , itd rightlimit and itd music are calculated from the phase of the appropriate transfer function . the centre frequencies , f , of each fft bin , k , are calculated from the fft size and sample rate . the music signal used for the examples below is samples n = 2049 : 2560 of “ bird on a wire ” after the music begins . with reference to fig1 there is shown ild music and itd music . with reference to fig1 ( left plot ) there is shown ild leftlimit and ild rightlimit . these ild and itd functions are part of the input to the mapping step in function block c ( reference numeral 21 ) in fig9 . the coherence between l source [ n ] and r source [ n ], which as mentioned above takes a value between 0 and 1 , is calculated from the power spectral densities of the two signals and their cross - power spectral density . the power spectral densities of l source [ n ] and r source [ n ] can be calculated in the frequency domain as the product of the spectrum with its complex conjugate as shown below : the cross - power spectral density of l source [ n ] and r source [ n ] can be calculated in the frequency domain as a product of l source [ k ] and the complex conjugate of r source [ k ], as shown below : the coherence can be calculated in the frequency domain by means of the following equation : c lr ⁡ [ f ] =  p lr 2  p ll · p rr c lr was calculated over 8 blocks in the examples shown here . c lr will be equal to 1 at all frequencies if l source [ n ]= r source [ n ]. if l source [ n ] and r source [ n ] are two independent random signals , then c lr will be close to 0 at all frequencies . the coherence between l source [ n ] and r source [ n ] for the block of music is shown in fig1 . this function block maps the interaural differences and coherence calculated in the functions a and b into a filter that will be used to extract the components of l source [ n ] and r source [ n ] that will be reproduced by the centre loudspeaker . the basic idea is that the contributions of the ild , itd and interchannel coherence functions to the overall filter are determined with respect to some threshold that is determined according to the angular range intended to be covered by the loudspeaker . in the following , the centre loudspeaker is assigned the angular range of − 10 to + 10 degrees . the ild thresholds are determined from the free field interaural transfer function for sources at − 10 and + 10 degrees . two different ways of calculating the contribution of ild to the final filter are briefly described below . in the first mapping approach , any frequency bins with a magnitude outside of the limits , as can be seen in fig1 , are attenuated . ideally the attenuation should be infinite . in practice , the attenuation is limited to a db , in the present example 30 db , to avoid artefacts from the filtering such as clicking . these artefacts will be commented further upon below . this type of mapping of ild to the filter is shown in fig1 . an alternative method is simply to use the negative absolute value of the magnitude difference between h iaff [ f ] for a source at 0 degrees and h iamusic [ f ] as the filter magnitude as shown in fig1 . in this way , the larger difference between h iamusic [ f ] and h iaff [ f ], the more h iamusic [ f ] is attenuated . there are no hard thresholds as in the method above and therefore some components will bleed into adjacent loudspeakers . as in the previous section , the itd thresholds are determined from the free field interaural transfer function for sources at − 10 and + 10 degrees , respectively . again , two methods for including the contribution of itd to the final filter are described below . the phase difference between h iaff [ f ] for a source at 0 degrees and h iamusic [ f ] is plotted with the itd thresholds for the centre loudspeaker in fig1 . the result of the first “ hard threshold ” mapping approach is the filter magnitude shown in fig1 . all frequency bins where the itd is outside of the threshold set by free field sources at − 10 and + 10 degrees , respectively , are in this example attenuated by 30 db . another approach is to calculate the attenuation at each frequency bin based on its percentage delay compared to free filed sources at − 30 and + 30 degrees , respectively . for example , if the maximum delay at some frequency was 16 samples and the itd for the block of music was 4 samples , its percentage of the total delay would be 25 %. the attenuation then could be 25 % of the total . that is , if the total attenuation allowed was 30 db , then the relevant frequency bin would be attenuated by 18 db . an example of the filter magnitude designed in this way is shown in fig2 . as intensity and time panning function best for coherent signals , the operation of the stereo to multi - mono conversion should preferably take the coherence between l source [ n ] and r source [ n ] into account . when these signals are completely incoherent , no signal should be sent to the centre channel . if the signals are completely coherent and there is no ild and itd , then ideally the entire contents of l source [ n ] and r source [ n ] should be sent to the centre loudspeaker and nothing should be sent to the left and right loudspeakers . the coherence is used in this implementation as a scaling factor and is described in the next section . the basic filter for the centre loudspeaker , h centre [ f ], is calculated as a product of the ild filter , itd filter and coherence formulated in the equation below . it is important to note that this is a linear phase filter — the imaginary part of each frequency bin is set to 0 as it is not desired to introduce phase shifts into the music . h center [ f ] = ildmap centre [ f ] · itdmap centre [ f ]· c lr [ f ] the result is a filter with a magnitude like that shown in fig2 . h centre [ f ] is updated for every block , i . e . it is a time varying filter . this type of filter introduces distortion which can be audible if the discontinuities between blocks are too large . fig2 shows an example of such a case where discontinuities can be observed in a portion of a 50 hz sine wave around samples 400 and 900 . two means to reduce the distortion are applied in the present implementation . first across - frequency smoothing is applied to h centre [ f ]. this reduces the sharp changes in filter magnitude of adjacent frequency bins . this smoothing is implemented by replacing the magnitude of each frequency bin with the mean of the magnitudes ⅓ of an octave to either side of it resulting in the filter shown in fig2 . note that the scale of the y - axis is changed compared with fig2 . slew rate limiting is also applied to the magnitude of each frequency bin from one block to the next . fig2 shows h centre [ f ] for the present block and the previous block . magnitude differences of approximately 15 db can be seen around 1 khz and 10 khz . the magnitude of these differences will cause audible distortion that sounds like clicking . the slew rate limiting is implemented with a conditional logic statement , an example of which is given in the pseudo - code below . choosing the values of maximum positive and negative change is a trade - off between distortion and having a filter that reacts quickly enough to represent the most important time - varying nature of the relationship between l source [ n ] and r source [ n ]. the values were in this example determined empirically and 1 . 2 db was found to be acceptable . fig2 shows the change between h centre [ f ] for the present block and the previous block using this 1 . 2 db slew rate limit . consider again the regions around 1 khz and 10 khz . it is clear that only the differences up to the slew rate limit have been preserved . fig2 shows the same portion of a 50 hz sine wave where across - frequency - smoothing and slew rate limiting has been applied to the time varying filter . the discontinuities that were clearly visible in fig2 are greatly reduced . the fact that the gain of the filters has also changed at this frequency is also clear from the fact that the level of the sine wave has changed . as mentioned above there is a trade - off between accuracy representing the inter - channel relationships in the source material and avoiding artefacts from the time - varying filter . if fast - convolution is to be used , which is equivalent to circular convolution , the filters must be converted to their time - domain forms so that time - aliasing can be properly controlled ( this will be more thoroughly described below ). the inverse discrete fourier transform , abbreviated idft and given by the following equation and referred to as the fourier synthesis equation of h centre [ k ] yields its impulse response . as h center [ f ] is linear phase , h center [ n ] is an acausal finite impulse response ( fir ) filter , n samples long , which means that it precedes the first sample . this type of filter can be made causal by applying a delay of n / 2 samples as shown in fig2 . note that the filter is symmetrical about sample n / 2 + 1 . the tap values have been normalised for plotting purposes only . the time to convolve two sequences in the time domain is proportional to n 2 where n is the length of the longest sequence . whereas the time to convolve two sequences in the frequency domain , that is the product of their frequency responses , is proportional to nlogn . this means that for sequences longer than approximately 64 samples , frequency domain convolution is computationally more efficient and hence the phrase fast convolution . there is an important difference in the output of the two methods — frequency domain convolution is circular . the curve shown in heavy line in fig2 is the output sequence of the time domain convolution of the filter in fig2 , length n = 512 , with a 500 hz sine wave , length m = 512 . note the 256 sample pre - ringing that is a consequence of making causal the linear phase filter . in this case the output sequence is ( n + m )− 1 = 1023 samples long . the light curve shown in fig2 is the output sequence of fast convolution of the same filter and sine wave and is only 512 samples long . the samples that should come after sample 512 have been circularly shifted and added to samples 1 to 511 , which phenomenon is known as time - aliasing . time - aliasing can be avoided by zero padding the sequence before the fourier transform and that is the reason of returning to a time domain representation of the filters mentioned in the section about function block d above . the heavy curve in fig2 is the output sequence of the time domain convolution of the filter in fig2 , length n = 512 , with a 500 hz sine wave , length m = 1024 . in this case the output sequence is ( n + m )− 1 = 1535 samples long . the light curve in fig2 is the output sequence of fast convolution of the same filter zero padded to a length n = 1024 samples and sive wave still with length m = 1024 . here the output sequence is 1024 samples long , however , in contrast to the case above , the portion of the output sequence in the same position as the zero padding , samples 512 to 1024 , is identical to the output of the time domain convolution . saving this portion and repeating the process by shifting 512 samples ahead along the sine wave is called the overlap - save method of fast convolution and is equivalent to time domain convolution with the exception of the additional 256 sample delay making the total delay associated with the filtering process filter_delay = 512 samples . reference is made to oppenheim and schafer [ 1999 , p . 587 ] for a thorough explanation of this technique . the signal to be reproduced by the centre loudspeaker , c output [ n ], is calculated using the following equations : the signals to be reproduced by the left and right loudspeakers , respectively , are then calculated by subtracting c output [ h ] from l source [ n ] and r source [ n ], respectively , as shown in the equation below . note that l source [ n ] and r source [ n ] are delayed to account for the filter delay filter_delay . l output [ n ] = z − filter _ delay · l source [ n ] − l filtered [ n ] r output [ n ] = z − filter _ delay · r source [ n ] − r filtered [ n ] in the special case where r source [ n ]=− l source [ n ], the signals are negatively correlated , and it is easy to show that all the output signals will be zero . in this case the absolute value of the phase of the cross - power spectral density , p lr [ k ], will be equal to π ∀ k and the coherence , c lr [ k ], will be equal to 1 ∀ k . the conditional statement in the pseudo - code below is applied to ensure the l output [ n ]= l source [ n ], r output [ n ]=− l source [ n ] and c output [ h ]= 0 . also in the case of silence on either l source [ n ] or r source [ n ], then c lr [ k ] should be zero . however , there can be numerical problems that prevent this from happening . in the present implementation , if the value of either p ll [ k ] or p rr [ k ] falls below − 140 db , then c lr [ k ] is set to zero . albert s . bregman . auditory scene analysis . the mit press , cambridge , mass ., 1994 . søren bech . spatial aspects of reproduced sound in small rooms . j . acoust . soc . am ., 103 : 434 - 445 , 1998 . jens blauert . spatial hearing . mit press , cambridge , mass ., 1994 . d . hammershøi and h . møller . sound transmission to and within the human ear canal . j . acoust . soc . am ., 100 ( 1 ); 408 - 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