Patent Application: US-84269904-A

Abstract:
a method for canceling interference at a wireless code division multiple access communication receiver is provided . the wireless cdma communication system receiver receives a stream of chips generated by spreading data symbols formed by grouping bits of information at a wireless cdma communication transmitter which are broadcast at a certain chip - rate . the received chips are de - spread and symbols pertaining to respective users are reconstructed . the method includes formatting the stream of chips into blocks of chips , and performing an iterative block decision feedback equalization in a frequency domain at the chip - rate of the broadcast stream of chips to remove inter - symbol interference by defining a transfer function . the transfer function is defined based upon iteration cycles as a function of data detected in a preceding iteration cycle . the chips generated are interleaved by spreading each data symbol being transmitted before broadcasting the stream of interleaved chips in distinct blocks of chips .

Description:
for ease of explanation the ensuing description is divided in the following sections : 1 ) the system model — which illustrates an embodiment of the system and the data transmission format that is used for an fd implementation of the receiver and the interleaving that is implemented at a chip level ; 2 ) interference cancellation and dfe — which describes the fd implementation of both sic and pic and the iterative block dfe detector ; and 3 ) design methods — which illustrates the manner in which the filter coefficients of the dfe are defined . system model . in terms of notation , a signal is denoted with a lowercase letter , while its dft is denoted with the corresponding uppercase letter . vectors and matrices are denoted with letters , and * denotes complex conjugate . in a wideband cdma communication system , where k users are transmitting simultaneously , codes with spreading factor n s are used . for the user k , the data signal d ( k ) ( n ), at rate 1 / t , is spread with the code c ( k ) ( m ), m = 0 , 1 , . . . , n s − 1 to obtain the data sequence ( chips ) q ( k ) ( m + nn s )= c ( k ) ( m ) d ( k ) ( n ), k = 0 , 1 , . . . , k − 1 , ( 1 ) having rate n s / t . after spreading , chips are interleaved as detailed below to obtain the signal s ′( k ) ( n ). for transmission , s ′( k ) ( n ) is modulated with the pulse - shaping transmit filter . in a wideband uplink transmission , for each user the channel can be modeled as the sum of delayed paths with different attenuations and phases . at the base station , the received signal is filtered by a filter matched to the transmit pulse shape and then sampled at the rate 1 /( tn s ), [ 2 ]. by denoting with h ( k ) ( q ), q = 0 , 1 , . . . , n q − 1 , the discrete - time base band equivalent impulse response of the channel for the user k , including pulse shaping and matched filtering , the received discrete - time signal comprises the sum of the signals of all users , i . e ., r ′ ⁡ ( l ) = ∑ k = 0 k - 1 ⁢ r ′ ⁡ ( k ) ⁡ ( l ) + w ⁡ ( l ) = ∑ k = 0 k - 1 ⁢ ∑ q = 0 n q - 1 ⁢ h ( k ) ⁡ ( q ) ⁢ s ′ ⁡ ( k ) ⁡ ( l - q ) + w ⁡ ( l ) , ( 2 ) where w ( λ ) is the complex additive white gaussian noise ( awgn ) term , having zero mean and variance n 0 / 2 per dimension . if despreading with the code of the user k is directly applied to r ′( λ ), the obtained signal will be affected by mai and isi . therefore , as already mentioned in the introduction , use of special and generally more complex receivers becomes unavoidable . according to the invention , a receiver is provided that includes joint ic and equalization functions , and wherein filtering is performed in the fd . data transmission format . to implement filter operations as a product of dfts on sampled blocks of size p , the convolution between the channel impulse response and the transmitted data signal must be circular on blocks of size p . two methods for forcing the circularity that have been proposed recently are briefly discussed below . in [ 14 ] it has been proposed to append to each block of m information data s ′( k ) ( m )=[ s ′( k ) ( mm ), s ′( k ) ( mm + 1 ), . . . , s ′( k ) ( mm + m − 1 )] ( 3 ) a known pn sequence [ p ( 0 ), p ( 1 ), . . . , p ( l − 1 )]. the extended data block of length p = m + l is given by s ( k ) ⁡ ( m ) = ⁢ [ s ( k ) ⁡ ( mp ) , s ( k ) ⁡ ( mp + 1 ) , … ⁢ , s ( k ) ⁡ ( mp + p - 1 ) ] = ⁢ [ s ′ ⁡ ( k ) ⁡ ( mm ) , s ′ ⁡ ( k ) ⁡ ( mm + 1 ) , … ⁢ , s ′ ⁡ ( k ) ⁡ ( mm + m - 1 ) , ⁢ p ⁡ ( 0 ) , p ⁡ ( 1 ) , … ⁢ , p ⁡ ( l - 1 ) ] . ( 4 ) this extension is denoted as zero - padding in the case p ( n )= 0 , n = 0 , 1 , . . . , l − 1 , [ 16 ]. note that an additional pn extension is required before the first data block . another technique that forces the convolution to be circular is the cyclic - prefix ( cp ) extension , where the first l symbols coincide with the last symbols of the extended blocks , i . e ., s ( k ) ( m )=[ s ′( k ) ( mm ), s ′( k ) ( mm + 1 ), . . . , s ′( k ) ( mm + m − 1 ), s ′( k ) ( mm ), . . . , s ′( k ) ( mm + l − 1 )]. ( 5 ) here the convolution is circular on blocks of size p = m . such a cp transmission has been proposed for use in multicarrier communications [ 15 ] and has also been proposed for single carrier transmissions with linear fd equalization , [ 17 ], [ 18 ]. note that , while according to the dfe technique proposed in [ 14 ], where the fb operates in the time domain , only the pn - extension method may be used . in the cp method of [ 15 ], [ 17 ] and [ 18 ] there is no such limitation . however , the pn - extension method yields a better performance than the cp method because it implies a reduced error propagation phenomena in the block dfe . for the same bandwidth efficiency , the cp method has a reduced complexity since dfts are performed on blocks of size m instead of p . in the following description the pn - extension method will be considered for illustrative purposes . however , variations of the implementation of this invention for the case of a cp method are straightforward and will be immediately recognized by a skilled person . to simplify the notation , the block index m will be omitted since all operations are performed on blocks of p samples of the received signal . chip interleaving . in a dfe , the use of past detected data produces an error propagation phenomena . the impact of errors is particularly significant because of cdma transmission and the fd implementation of the dfe . in fact , the error on one data symbol is propagated by the spreading on an entire cdma symbol ( i . e ., n s chips ), thus generating a burst of errors . it has been found that error propagation may be significantly reduced if the chips are interleaved before transmission . because data must be detected at each block to allow dfe , the interleaver must operate separately on each block of transmitted chips . in particular , a row - column block interleaver where the m information chips are written in a matrix column - wise and read row - wise [ 2 ] is a preferred embodiment . the ensuing description will assume the use of such an interleaver . fig2 shows an example of chip interleaving with a row - column interleaver of size w × q , with wq = m . note that the superscript ( k ) has been omitted for ease of notation . interference cancellation and dfe . the most common structure for wideband cdma transmissions comprises a rake , i . e ., a filter matched to the channel impulse response , followed by a despreader and a detector . in our fd implementation , the matched filter operation yields the fd signal y p ( k ) = h p ( k ) * r p , p = 0 , 1 , . . . , p − 1 ( 6 ) note that in an fd implementation , no reduction of complexity is achieved by considering only a subset of the coefficients ( fingers ) of the channel impulse response . in a traditional basic receiver , after the matched filter , the vector y p ( k ) = [ y 0 ( k ) , y 1 ( k ) , … ⁢ , y p - 1 ( k ) ] would be transformed into the time domain by an idft . the signal is then despread . lastly , detection is performed . in an advanced receiver , the impact of mai can be reduced by applying an interference cancellation scheme where , starting with an initial tentative decision of symbols of one or more users , their interference contribution is generated and subtracted from the receiver signal . after cancellation and despreading have been performed , a new detection is applied . of course , the cancellation and detection procedures may be iterated a few times to increase the reliability of data . serial interference cancellation ( sic ). in sic , the ic is performed serially for each user and its fd implementation ( fd - sic ) is shown in fig3 . the received signal is first divided into blocks of size p by a serial to parallel converter ( s / p ), and each block is transformed into the fd by means of a dft of size p , to obtain r p ⁡ ( m ) = ∑ l = 0 p - 1 ⁢ ⅇ - j ⁢ ⁢ 2 ⁢ π ⁢ ⁢ lp / p ⁢ r ⁡ ( mp + l ) , p = 0 , 1 , … ⁢ ⁢ p - 1 . ( 7 ) similarly , let &# 39 ; s denote with s ( k ) ( m ), w ( m ) and h ( k ) the dfts of s ( k ) ( m ), w ( m ) and h ( k ) , respectively . the received signal can be written in the frequency - domain as r p ⁡ ( m ) = ∑ k = 0 k - 1 ⁢ h p ( k ) ⁢ s p ( k ) ⁡ ( m ) + w p ⁡ ( m ) . ( 8 ) the following operations perform a multistage detection and ic of each user signal . at the first stage , r is used as input of the block interference 0 , which performs data detection of the user k = 0 and the subsequent interference generation which is canceled from the input of the next stage . in general , in the basic sic receiver , the interference k block includes the matched filter h ( k ) * ( or rake ), the despreader and the detector which generates the signal d ( k ) . to obtain the interference contribution in the fd , v ( k ) , d ( k ) is first spread to form s ′( k ) which in turn yields the augmented block s ( k ) by appending the pn sequence ( see ( 5 )). s ( k ) is then transformed into the fd to obtain the vector s ( k ) , which is multiplied by the frequency response of the corresponding channel , i . e ., v p ( k ) = h p ( k ) ŝ p ( k ) , p = 0 , 1 , . . . , p − 1 . ( 9 ) a more detailed description of the interference k block will be presented in the following . with regards to fig3 , each stage of sic includes cancellation of the interference contribution of previously detected users . hence , the input of the interference k block is given by x ( k ) = r - ∑ l = 0 k - 1 ⁢ v ( l ) = x ( k - 1 ) - v ( k - 1 ) , k = 1 , 2 , … ⁢ , k - 1 . ( 10 ) the order of cancellation of the various users is varied according to the power of the received user signals and their interference contribution to other users , [ 19 ]. in fact , the effect of mai on the x ( k ) signal diminishes from the first to the last stage , and for equal user bit error rates ( ber ) proper power control must be implemented [ 20 ]. parallel interference cancellation ( pic ). in pic , ic is performed in parallel for all users and its fd implementation ( fd - pic ) is shown in fig6 . as in sic , also here the interference k block performs detection and generation of the interference contribution of user k . the following operations perform iteratively detection and ic of user signals for i times . reference is made to pic iterations in the parallel ic process to avoid confusion with term iteration which is reserved for the equalization - detection process . at the first pic iteration ( i = 1 ), all feedback signals of fig6 are not active and the received signal is applied to a bank of interference k blocks , performing parallel detection of all user data signals and generation of mai . at pic iterations i = 2 , 3 , . . . , i , the input to each interference k block is obtained by canceling from the received signal the mai due to all other users , as shown in the efficient implementation of fig6 . in this case , the signal at the input of the interference k block is x ( k ) = r - ∑ l = 0 , l ≠ k k - 1 ⁢ v ( l ) , k = 0 , 1 , … ⁢ , k - 1 . ( 11 ) as an alternative , cancellation could include also isi [ 21 ], and in this case x p ( k ) = r p - [ h p ( k ) - ⅇ - j ⁢ ⁢ 2 ⁢ π ⁢ ⁢ p ⁢ ⁢ δ l / p ⁢ h ( k ) ⁡ ( δ k ) ] ⁢ s ^ p ( k ) - ∑ l = 0 , l ≠ k k - 1 ⁢ v p ( k ) , ⁢ k = 0 , 1 , … ⁢ , k - 1 , ( 12 ) p = 0 , 1 , . . . , p − 1 , where δ λ is the delay corresponding to the coefficient with the largest amplitude of the channel impulse response of user λ . note that in the scheme of fig6 the ic removes the contribution of other users . to take into account the reliability of the tentative decision , in [ 22 ] it is proposed to weight the signals before cancellation and obtain a trade - off between cancellation of interference and insertion of a disturbance due to decision errors . this weighting can be easily accomplished in the fd - pic scheme by multiplying v ( k ) with an appropriate gain before cancellation . for a coded transmission , since all operations are performed on blocks of cdma symbols , the detected signal can be also decoded before fd - pic is performed , [ 11 ]. equalization and interference generation . both fd - pic and fd - sic include the interference k blocks that will be described here in more detail . while the outer ic is used to reduce mai , the purpose of the interference k block is to regenerate the interference due to user k in the received signal . indeed , this signal is affected by isi , and for better detection ( and hence regeneration ), a dfe is used . in other words , the traditional matched filter ( or rake ) used in prior art architectures is substituted according to an aspect of the invention with an iterative dfe to reduce isi . contrary to the dfe proposed in [ 9 ], where the fb signal was designed to cancel only the interference among contiguous cdma symbols , according to the approach of the invention the dfe operates at the chip - rate , hence it is more effective in equalizing . the functional diagram of each interference k block is shown in fig7 . note that , due to the cdma transmission format , a time - domain dfe can not be applied directly on the received signal , since despreading must be performed on blocks of chips before the fb signal is available . according to the invention , an implementation of a block dfe operating in the fd is described , wherein equalization and detection are iterated v times on the input signal x ( k ) . in view of the fact that all fd signals are already available , the dfe scheme reduces to the simple operations shown in fig6 . at the iteration l , the feed - forward filter is implemented in the fd by the element - wise complex multiplication of x ( k ) with the vector of the ff filter coefficients c ( k , l ) = [ c 0 ( k , l ) , c 1 ( k , l ) , … ⁢ , c p - 1 ( k , l ) ] , l = 1 , 2 , … ⁢ , v . b ( k , l ) = [ b 0 ( k , l ) , b 1 ( k , l ) , … ⁢ , b p - 1 ( k , l ) ] . when the dfe is first applied ( l = 1 ) to the received vector signal , no previous decision is available and the feedback signal is not active , i . e ., b p ( k , l ) = 0 , p = 0 , 1 , . . . , p − 1 . in this case the dfe k becomes a linear equalizer and the resulting vector signal has elements y p ( k , l ) = x p ( k ) c p ( k , l ) , p = 0 , 1 , . . . , p − 1 . ( 13 ) y ( k , l ) is then transformed into the time domain by idft . after de - spreading ( ds ), detection ( det ) yields the tentative decision vector d ( k , l ) . the detailed description of the various blocks is shown in fig7 , where parallel to serial conversion is denoted with p / s . note that after the idft of the vector signal y ( k , l ) the last l symbols are discarded before de - interleaving ( deint ) and de - spreading is performed , and detection follows . there are also linear ic schemes that do not include detection [ 24 ]. a new equalization iteration can now be applied . first , the detected data block d ( k , l ) is regenerated by spreading ( sp ), interleaving ( int ) and pn insertion . the whole block is transformed in the fd by dft to obtain the data vector s ( k , l ) . then , equalization of the input signal is performed which now includes the fb coefficients b p ( k , 2 ) , p = 0 , 1 , . . . , p − 1 in the dfe , as shown in fig6 . in general , from the second iteration on ( i . e . l & gt ; 1 ) the dfe k generates the vector signal y ( k , l ) with elements y p ( k , l ) = x p ( k ) c p ( k , l ) + b p ( k , l ) ŝ p ( k , l − 1 ) , p = 0 , 1 , . . . , p − 1 . ( 14 ) an algorithm for the iterative design of both ff and fb filters according to the minimum mse criterion is described in the following the equalization and detection processes are iterated v times , until a reliable decision vector s ( k , v ) is available which is multiplied by the corresponding channel frequency response to obtain the interference contribution in the fd that must be canceled from the received signal v p ( k ) = h p ( k ) ŝ p ( k , v ) , p = 0 , 1 , . . . , p − 1 . ( 15 ) design methods . for the design of the dfe filters , minimization of the mse at the detection point is preferred , which at iteration l = 1 , 2 , . . . , v , can be written in the fd as in [ 25 ] j ( k , l ) = 1 p 2 ⁢ ∑ p = 0 p - 1 ⁢ e ⁡ [  c p ( k , l ) ⁢ x p ( k ) + b p ( k , l ) ⁢ s ^ p ( k , l - 1 ) - s p ( k )  2 ] . ( 17 ) by assuming that both transmitted and detected data are random variables , with zero - mean , and statistically independent from the noise , the expectation in ( 17 ) with respect to data and noise signals , yields the mse j ( k , l ) = ⁢ 1 p 2 ⁢ ∑ p = 0 p - 1 ⁢ ⁢  c p ( k , l )  2 ⁢ m w +  c p ( k , l ) ⁢ h p ( k ) - 1  2 ⁢ m s +  b p ( k , l )  2 ⁢ m s + ⁢ 2 ⁢ ⁢ re [ b p ( k , l ) * ⁡ ( c p ( k , l ) ⁢ h p ( k ) - 1 ) ⁢ r ( k , l ) ] , ( 18 ) where m w = pn 0 is the noise power in the fd , m s is the power of each element of s ( k ) , and r ( k , l ) is the correlation between the transmitted data and the detected data at the previous iteration , [ 23 ] r ( k , l ) = e [ d ( k ) ( n ) { circumflex over ( d )} ( k , l − 1 ) *( n )]. ( 19 ) the correlation depends on the channel and on the noise level and it must be estimated at the receiver . here we summarize a method , whose details are given in [ 25 ], where an estimate of r ( k , l ) is obtained by x ( k ) and the fd detected signal at the previous iteration , s ( k , l − 1 ) , r ^ ( k , l ) = η ⁢ ∑ p = 0 p - 1 ⁢ ⁢ x p ( k ) h p ( k ) ⁢ s ^ p ( k , l - 1 ) * , ( 20 ) with η being a correction factor ( η & lt ; 1 ) to reduce the dfe error propagation phenomena . in this design method , since the reliability of the detected signal at the feedback input is increasing with the number of iterations , the filters will be different at the various iterations . to derive the filters that minimize ( 18 ), the constraint that the fb filter removes pre - and post - cursors , but does not remove the desired component , is imposed . in other words , it must be ∑ p = 0 p - 1 ⁢ ⁢ b p ( k , l ) = 0 . ( 21 ) the application of the gradient method to minimize ( 18 ), with respect to the fb filter coefficients , which yields b p ( k , l ) = - r ^ ( k , l ) m s ⁡ [ h p ( k ) ⁢ c p ( k , l ) - γ ( k , l ) ] , p = 0 , 1 , … ⁢ , p - 1 , ⁢ where ( 22 ) γ ( k , l ) = ∑ p = 0 p - 1 ⁢ ⁢ h p ( k ) ⁢ c p ( k , l ) , ( 23 ) c p ( k , l ) = h p ( k ) * m w + m s ⁡ ( 1 -  r ^ ( k , l )  2 m s 2 ) ⁢  h p ( k )  2 . ( 24 ) special attention should be paid to the first iteration of the dfe - detector ( l = 1 ), which yields different approaches , according to the adopted ic scheme . fd - sic . at each sic stage ( i . e ., user k detection ), for l = 1 no tentative decision is available , hence { circumflex over ( r )} ( k , l ) = 0 , b ( k , l ) = 0 and the ff filter is a linear minimum mse equalizer . fd - pic . at the first pic iteration ( i = 1 ) no tentative decision is available at the first dfe iteration , hence { circumflex over ( r )} ( k , l ) = 0 and b ( k , l ) = 0 , for k = 0 , 1 , . . . , k − 1 . at the next pic iterations ( i & gt ; 1 ), a tentative decision of the previous pic iteration is available also for l = 1 and a dfe can be used for each user . a significant indication of the improvement that the method of the present invention achieves may be gained by observing a comparison between the cumulative distribution function of the normalized power ζ with ( continuous line curve ) and without ( dashed line curve ) chip interleaving walsh codes with ns = 16 as shown in fig8 . h . holma and a . toskala , wcdma for umts : radio access for third generation mobile communications . new york : wiley , 2 ed ., 2000 . j . g . proakis , digital communications . new york : mcgraw hill , 1994 . s . verd &# 39 ; u , multiuser detection . cambridge , u . k . : cambridge univ . press , 1998 . a . klein , “ data detection algorithms specifically designed for the downlink of cdma mobile radio systems ,” in proc vehic . tech . conf . 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