Patent Application: US-47193802-A

Abstract:
a radio transmission method and a wireless transmission system comprising multiple transmitter and receiver antennae simultaneously operating within the same frequency range and space - time encoders and decoders . data transmission and channel matrix measurement can take place synchronously . characteristic pilot signals that are unequivocally recognizable are superimposed with low noise on the data subsignals in the transmitter and are used in the receiver for the analog measurement of the channel matrix in a weighting unit and for the analog recovery of the transmitted data subsignals from the received signals . the measured analog values are digitally processed in a signal processor . the weightings thus determined are then summed in an analog signal processing unit .

Description:
reference will now be made in detail to the preferred embodiments of the present invention , examples of which are illustrated in the accompanying drawings , wherein like reference numerals refer to like elements throughout . fig1 depicts the structure of a multiple input / multiple output system mimo as is known from the state of the art , with the relation of a transmitter tx at the transmission side to a receiver rx at the receiving side over a multipath transmission channel in an indoor space . fig2 depicts the basic scheme of the transmission method and system ts in accordance with the invention which will hereafter be explained in greater detail . the new concept of channel matrix measuring will be explained hereafter as the essential component of the wireless transmission method in accordance with the invention . basically , a modulated weak auxiliary channel carrier is added to each transmission path . hence , a pilot signal characteristic for each transmission path is superimposed on each data signal . amplitude and phase of the auxiliary channel carrier associated with each transmission path are then measured on each receiving path . the frequency of the auxiliary channel carrier in particular is identical to the frequency of the carrier for the data transmission ( identical channel operation ). the concept is exemplarily depicted in fig3 in connection with a transmitter tx which encodes the transmission signals by means of binary phase modulation ( binary phase shift keying — bpsk ) as the data modulation format , with two transmission paths or antennae s 1 , s 2 ( s i , i = 1 . . . n , n = 2 ). initially the incoming data stream data - in is divided in a space - time encoder ste to yield two parallel data subsignals d 1 ( t ), d 2 ( t ). in each transmission path s 1 , s 2 their mean is assumed to be balanced , i . e . the amplitude averaged over a large number of symbols equals nil . it is additionally to be assumed that the chronological sequence of the data subsignals d 1 ( t ), d 2 ( t ) is statistically distributed in a purely random manner , which can be achieved by “ scrambling ”. to each of the data subsignals d 1 ( t ), d 2 ( t ) a different pilot signal p 1 ( t ), p 2 ( t ) is added , the pilot sequences pn 1 , pn 2 ( pseudo noise pn ) of which are generated by an associated pseudo random generator ( generator ) for binary symbol sequences ( pseudo random binary sequence prbs ) and are modulated onto an mean frequency if , as a carrier for the data subsignals d 1 ( t ), d 2 ( t ). the pilot signals p 1 ( t ), p 2 ( t ) are very weak so that they do not interfere with the data subsignals d 1 ( t ), d 2 ( t ). the factor k which describes the amplitude ratio between the superimposed signals is significantly smaller than 1 ( k & lt ;& lt ; 1 ). the superimposed signals are then raised to the transmission band by means of a local frequency generator lo , guided through band filters fi and amplifiers a , and are transmitted into the indoor space as time - dependent transmission signals s 1 ( t ), s 2 ( t ). for unequivocally distinguishing the marking pilot sequences pn i of different transmission paths s 1 in the receiver rx , it is important to use binary symbol sequences prbs of good correlation properties . in addition , relatively long code sequences ( length l =& gt ; 2 14 − 1 = 16 , 383 ) are required to distinguish the relatively weak pilot signals p 1 ( t ) from the relatively strong transmitted data signals d i ( t ) in correlation circuits . such long binary symbol sequences prbs may be generated by multiple - feedback shift registers . the actual feedback ratios may be derived from the generally known table of the non - shortenable binary polynomials [ 7 ]. there are many possible feedback ratios which generate different sequences of the same maximum length l ; but in general the cross correlation of an arbitrary pair of these sequences is not optimal . a double shift register sr which utilizes a preferred pair of sequences is known from gold [ 8 ]. a whole family of such sequence pairs of good auto and correlation properties can be obtained with the gold sequence generator . one way of determining the preferred pair is shown in [ 9 ]. it is determined by l = 16 , 383 . the two preferred polynomials f 1 ( x )= x 14 + x 10 + x 6 + x + 1 and f 65 ( x )= x 14 + x 13 + x 11 + x 10 + x 9 + x 8 + x 6 + x 3 + x 2 + x 1 + 1 may be found in the table [ 7 ]. the exponents of the polynomials identify tabs in the shift registers sr which are summed by modulo 2 and then feedback to the input of the register . it is to be noted that while they cause a delay , different initial values in the shift register generate identical sequences . “ modulo ” is a mathematical direction for determining the whole number remainder of the division of two whole numbers . “ modulo 2 ” means that an arbitrary real number is rounded up or down , divided by 2 , and the remainder is issued as a result of either “ 1 ” or “ 0 ”. in practice , this operation is performed by means of binary xor gates . the gold sequence generator thus set up is depicted in fig4 . the output of the lower shift register sr 2 is delayed by several chip cycles t ( chip = duration of bit ) and is then added to the output of the upper shift register sr 1 . here , the tabs are 1 , 6 , 10 and 14 . the tabs in the lower shift register sr 2 are 1 , 2 , 3 , 6 , 8 , 9 , 10 , 11 , 13 , and 14 . by changing the delay between the preferred sequences n & lt ; l + 2 different gold sequences (“ gold codes ”) can be obtained . by computer simulation it was tested that with the arrangement shown in fig3 , gold sequences are obtained which satisfy the following auto ( pacf [ i ] and cross correlation functions ( pccf [ i ]: using pilot signals p 1 ( t ) of identical symbol rates and , especially , of a modulation format of a similar spectral power distribution as in the data pccf ⁡ [ i ] = { - 1 ⁢ : ⁢ 1 = 0 ε ⁢ { 255 · - 1 , - 257 } ⁢ : ⁢ i ≠ 0 signals d 1 ( t ), d 2 ( t ) ( or in the incoming data stream data - in ) is a suitable method of taking into consideration the frequency selective fading in the channel which is usually very strong at an increasing transmission band width . because of its simplicity , the bpsk technique is preferably used for the channel estimation . it is to be notes that many higher order modulation techniques ( qpsk , 8 - psk , 16 - qam , 32 - qam , . . . ) have the same power distribution as the bpsk technique , provided the same symbol rate is utilized . for that reason , a bpsk - based pilot signal modulation can be combined , for instance , with data modulation based on 16 - qam . the sequence marked pilot signals p 1 ( t ) generate a kind of pseudo accidental noise which is added to the data subsignals d i ( t ). this reduces the transmission power somewhat , yet if the power of the pilot signals p i ( t ) amounts to only 1 % to 10 % of the partial data streams d i ( t ), its effect on the data transmission is negligibly small . the interference effect may be kept even smaller by subtracting the pilot signals p i ( t ) from the reconstructed transmission signals s i ( t ) ( see fig2 ). fig5 displays a receiver rx . at the left portion , there is shown a separation unit seu for the separation into signal and monitor paths , the center portion shows an analog signal combining unit asu and the right portion shows a data recovery unit dru . the receiver rx shown by way of example has three receiving paths or antennae e 1 , e 2 , e 3 ( e j , j = 1 . . . m , m = 3 ,). after band lowering by internal oscillators lo , each receiving signal e 1 ( t ), e 2 ( t ), e 3 ( t ) is divided at branches sp 1 , sp 2 , sp 3 to signal paths signal 1 , signal 2 , signal 3 and monitor paths m 1 ( t ), m 2 ( t ), m 3 ( t ). the corresponding monitor signals m 1 ( t ), m 2 ( t ), m 3 ( t ) are needed for measuring ( estimating ) the channel matrix { h }, whereas the receiving signals proper e 1 ( t ), e 2 ( t ), e 3 ( t ) are further processed in an analog signal combining unit asu . fig6 depicts details of a weighting unit wu as a connecting link between the separation unit seu and the signal combining unit asu for estimating the channel matrix [ h ]. in each receiving path e 1 , e 2 , e 3 the monitoring signals m 1 ( t ), m 2 ( t ), m 3 ( t ) are distributed onto a of correlation circuits correlator 1 , correlator 2 ( see upper portion of fig6 ) the number of which corresponds to the number of transmission antennae s 1 , s 2 ( by way of example , two transmission antennae at three receiving antennae ) in the transmission system . the number of all correlation circuits correlator ij is the product of multiplying n transmission paths with m receiving paths ( each with index i or j ). by way of example , the lower portion of fig6 depicts the correlation circuit correlator 1 associated with the marking pilot sequence pn 1 of the first transmission antenna s 1 . in synchronism with binary symbol sequence prbs of the pilot signal pn 1 in the received data signal e 1 ( t ), a prbs generator in the receiver rx generates the same symbol sequence prbs in order to generate the same pilot sequence pn1 as on the transmission side . an output of the same phase ( o °) and an output phase shifted by 90 ° of the internal if1 generator are then each modulated with the pilot sequence pn1 and each of the two resultant signals is multiplied by the monitor signal m 1 ( t ). in this connection , it is to be noted that the monitor signals m 1 ( t ) were mixed down to the subcarrier frequency if1 before the branch sp 1 , so the result of the multiplication is a complex valued base band signal . both superimposed signals are integrated during a full period τ of the symbol sequence prbs ( 16 , 383 chip clock cycles ). if during data transmission , the chip rate is 2 . 5 mhz for instance , a period of 6 . 5 ms is needed for each new measurement of the channel matrix [ h ]. this correctly assumes that the conditions for the multi - path transmission channel in the indoor space at 5 ghz do not significantly change during the duration of this period because of movements of objects at velocities & lt ; 1 m / s . the extraction of the pilot signal will now be described in greater detail . the monitor signal m 1 ( t ) arriving at the correlation circuit correlator 1 , which identical to the receiving signal e 1 ( t ), is a linear combination of two independent data subsignals d j ( t ) and two pilot signals p j ( t ) in the embodiment , this is the sum of four bpsk - coded signals on the same carrier frequency ω e 1 ⁡ ( t ) = 1 2 ⁢ ∑ j = 1 4 ⁢ [ h ij ⁢ s j ⁢ ⅇ j ⁡ ( ω ⁢ ⁢ x + π ⁢ ⁢ x j ⁡ ( t ) ) + c . c ] wherein x j ( t ) is the unipolar { 0 ; 1 } representation of the data or pilot signals and h 1j = h 1 ( j − 2 ) for j & gt ; 2 and c . c . is the abbreviation for the complex conjugate signal . the summation according ( 6 ) runs from j = 1 to 4 , because the calculation described here for data and pilot signals is now identical . this assumes that one and the same coherent frequency carrier is used for all transmission paths . the two signals measured behind the integration may be described as follows . in this context the i - signal may be interpreted as the real component and the q - signal as the imaginary component of the analog measuring voltage . the phase equal i - signal of the first transmission path s 1 may be found by insertion of ( 6 ) into ( 7 ). a similar equation results for the q - signal . the high frequency portion ( left bracket ) in ( 9 ) may be ignored since the integration functions in the manner of a low pass . if the properties of the channel do not change during period τ ( h ij = constant ), the result is integrals structured as c = ∫ 0 τ ⁢ ⅆ t ⁢ ⁢ ⅇ jx ⁡ [ x j ⁡ ( t ) - p 1 ⁡ ( t ) ] , which state whether or no a signal contributes to the measured i and q values . if x j ( t ) is one of the prbs sequences used for the time expansion of the pilot signal p j ( t ), then c is either equal to periodic auto correlation function ( pacf ( 0 ), j ≠ 1 ) or to the cross - correlation function ( pccf ( 0 ), j ≠ 1 ). the prerequisite is , however , that the pilot sequences , which unfold an n - dimensional vector space , be exactly orthogonal and synchronous to each other ĉ = tδ 1j , δ 1j being the kronecker symbol of all natural pairs ij for describing matrices of values 1 for i = j and 0 for i ≠ j . for the gold sequences here selected , these conditions are satisfied by a very good approximation . if x j ( t ) is an accidentally balanced data sequence , the resultant integral is c = 0 at an infinite period of the sequence . for that reason , the accidental data contributions will hereafter be ignored . in that case , the i and q signals are reduced to : i = s 1 4 ⁢ re ⁡ ( h 11 ) ⁢ ⁢ q = s 1 4 ⁢ im ⁢ ⁢ h 11 ) if the pilot signals in the transmitter are of the same amplitude , the amplitude ⁢ φ 11 = arctan ⁡ ( q i ) of the element h 11 of the channel matrix [ h ] may be measured . the definition of the other matrix elements is carried out analogously thereto . it is possible , however , with finite sequences sufficiently to suppress the contributions of the data signal to the measured i and q values as well , since they are not correlated to the pilot sequences . the accidental data signals may then be regarded as a strong noise superimposed on the pilot signals . in the following example , a pilot signal is assumed to have 1 % to 10 % of the energy of a data signal . since the data signals are correlated neither accidentally nor amongst each other , the energy of 23 db is above that of the pilot signal . following correlation , the pilot signal is amplified by about 10 log ( 16383 )= 42 db , by the gain of expansion , so that with 19 db , it now contributes more energy to the i and q signals ( or with the 9 - fold amplitude ) compared to the data signals . it is to be noted that the ratio of energy between partial data and pilot signals must be carefully set ( defining the rice factor k & lt ;& lt ; 1 ) for low cross - talk and high accuracy during channel measurement . the optimum relationship may be determined by measuring the bit error or by simulation of the entire system . it has already been mention above that while an alternating transmission of the pilot signals the spectral efficiency is again reduced somewhat , but that the amplitude of the pilot signals may be raised ( k → 1 ) to compensate . circuits similar to the ones shown in fig5 and 6 are used for separately measuring all elements h 11 . behind the integrators in fig6 the analog values are read into a signal processor dsp which calculates the weightings for the analog signal combining unit asu . it will now be shown how a matching weighting matrix ( i ] is obtained for the signal combining unit asu with the measured channel matrix [ h ]. by multiplying the vector description according to ( 2 ) from the left by the weighting matrix [ i ], the result will be s ′=[ i ][ h ] s +[ w ] n = s +[ i ] n . if noise is absent ( n = 0 ) it becomes clear that the weighting matrix [ i ] is the pseudo inverse of the channel matrix [ h ] which is defined by it is to be noted that ( 14 ) makes available m × n equations for the m × n elements of the weighting matrix [ i ], and that there are ( m − n )× n degrees of freedom in the calculation of the weighting matrix [ i ]. if noise is taken into consideration , the selection of an arbitrary weighting matrix [ i ] satisfying ( 14 ), may lead to significant errors in the signal recovery . this is clearly expressed by the additional noise term in s ′= s +[ i ] n which depends upon the actual settings of the free parameters in the weighting matrix [ i ]. in order to achieve an optimum data transmission , the effect of the noise term [ i ] n must be minimized . this task is accomplished , for instance , by the moore - penrose - pseudo inverses . the effect of this special selection of inverses will now be explained . it is to be noted that ( 2 ) describes a transformation between the n - dimensional transmission vector space σ ( n ) and the m - dimensional receiving vector space ψ ( m ) . in the absence of noise , the receiving vector e will thus be positioned in a hyperplane in the receiving vector space ψ ( m ) which is unfolded by the column vectors c i of the channel matrix [ h ]=[ c i , . . . c n ] ( see fig7 ). in the presence of noise the receiving vector e will normally be positioned outside of this plane . the most probable receiving signal ē 11 in the hyperplane may then be found by projection of the receiving vector e onto this plane , provided the noise in all receiving paths is of the same characteristic ( geometric projection technique ). if the normal noise contribution n i is removed , a one - to - ne relationship will result between the point in the plane and the transmission vector s . it is known from higher linear algebra that the moor - penrose pseudo inverses which may be found by means of a singular value breakdown of the complex valued channel matrix , carries out this sought projection . the noise dependent deviation from the original signal is defined correspondingly and may be interpreted as the retransformation of the noise ( 14 ) in the plane . if no information regarding the noise characteristic is available , this method which minimizes noise in the receiving space ψ ( m ) , is optimally suitable for the recovery of the transmission vectors s . usually , the noise behavior in the receiver is known , however , so that the error in the transmission vector space σ ( n ) may be further reduced with appropriate processes . minimizing the mean square deviation ( minimum means square estimation mmse ) causes small damage only ( 14 ); but it can weigh the residual error during data recovery ( cross - talk ) relative to the noise proportion . in this manner , it is possible to achieve improved efficiency by an algebraic approach which is hardly more difficult than the multiplication by the moor - penrose pseudo inverses . by contrast , the approach used by the known blast method uses a recursive interference extinction following the multiplication by the moore - penrose pseudo inverses . as a result of ( 14 ), this belated “ freeing from interference ” is , however , only necessary if there are errors in the measurement of the channel matrix as may be assumed in connection with the known blast method . in consequence of the resulting measurement error , a faulty pseudo inverse was here calculated which during reconstruction of the signals led to cross - talk between individual data paths . in the blast method , an improved channel estimation would require more time which would further reduce the spectral efficiency . the recursive suppression of interference in the blast method at least partially eliminates this undesired cross - talk and , hence , the consequences of a faulty channel measurement . by a significantly improved channel measurement it is thus possible to avoid the recursive error correction which is difficult to implement by hardware . by comparison with the very simple measuring method mentioned in ep a2 0 , 951 , 091 , the transmission system in accordance with the invention leads to a substantially higher accuracy by utilization of orthogonal pn sequences ( gold sequences ) so that complex interference suppression may be dispensed with . in this manner , the reconstruction of the partial transmission signal is substantially simplified by the improved method of measuring the channel matrix . only in this manner does the purely analog signal reconstruction described in the embodiment become practicable . the matrix calculation required for ( 19 ) is based upon measuring of a mean time value of the channel matrix [ h ]. data processing may thus be carried out in a digital signal processor dsp independently of the data transmission . aside from the required conversions at the input or output from analog to digital and vice versa ( a / d or d / a ), the digital signal processor dsp shown in fig6 depicts the necessary calculation steps for determining the individual elements h ij of the channel matrix [ h ] from the parameters i and q on the basis of which the pseudo inverses are calculated which in turn are passed along to the i - g modulators ig mod ij by way of the analog values i and q . in the analog signal combining unit asu the results of the calculation may now be used for executing the final steps of the hardware vector signal recovery , and in particular for the multiplication by the weightings i ij provided in an analog format by the i - q modulators iq mod ij and for the summing of the different contributions from all transmission antennae . the number of the required i - q modulators iq mod ij corresponds to the number of elements h ij of the channel matrix [ h ]. in addition to the correlation circuit for measuring the channel matrix , the analog signal combining unit asu in the center of fig5 constitutes a major component of the transmission system in accordance with the invention on the side of the multi - antenna receiver rx . since all operations affecting the data path are being realized by hardware , the throughput is not limited by the efficiency of the digital signal processor dsp . it is to be noted at this point that a hardware realization of this kind naturally may also be utilized by transmission systems other than one in accordance with the invention , especially there where data signals from a linear combining network have hitherto been processed exclusively in digital signal processors . among these are , in particular , so - called “ intelligent antenna arrays ” ( smart antennae ) which will find widespread use , for instance , for spatially separating individual participants at the base stations of future mobile wireless systems . in the concrete example , amplitude and phase of an mean frequency if 2 generated by a separate oscillator are adjusted by the digital signal processor dsp using conventional i - q modulators iq mod ij in accordance with every determined weighting i ij ( see right side of fig5 and asu in fig4 , here indicated by the expression “ i ij @ if 2 ). such modulators are known per se are usually used in digital radios for modulating the digital data onto a carrier . it is to be notes that the use of such a simple i - q modulator constitutes a novel way of weighting antennae in which a signal processor in the data paths becomes unnecessary . normally , a signal processor in the data path is indispensable in connection with intelligent antenna systems as those of the known blast method . the signals received on the mean frequency if , are now individually reduced to the intermediate frequencies if 3 = if 1 − if 2 so that the quasi - static amplitude and phase data of the complex weighting i ij of the mean frequency carrier if 2 is transferred to the received signals . the weighted signals on the mean frequency if 3 are then individually filtered in band passes fi and summed to the associated data branch . finally , the data are for into a space - time decoder std for recovery . in the example described above , the original spectral efficiency of a binary pulse layer modulation ( bpsk ) is multiplied by a number of two transmission antennae . assuming single side bands filters at the transmitter and considering that a band width 0 . 8 times the data rate is required in order reliably to detect the data in the base band , the spectral efficiency at bpsk is 1 . 25 bit per s and hz . for this reason , an efficiency of 2 . 5 bits per s and hz is realized in the example . the overall efficiency η total is increased to η total = n × η mod with more transmission antennae or different methods of higher order modulations in the separated partial data streams , wherein n is the number of partial data streams and η total is the spectral efficiency of the respective modulation of the partial data streams . at n = 8 and η mod = 5 ( for instance , in 16 - qam modulation ) may lead to a spectral efficiency η total of 40 bits per s and hz . assuming a natural channel band width of 2 mhz as in the above example , it will be possible at a chip rate of 2 . 5 mhz to transmit a data rate of 100 mbit / s . such a high data rate must be expected of a wireless lan , as has already been mentioned above . measured by the present hiperlan / 2 standard which utilizes a maximum spectral efficiency of 54 mbit / s and 20 mhz = 2 . 7 bit per s and hz , it is possible to increase the network capacity by a factor of almost 15 within the same spectral limits , with the transmission method in accordance with the invention and a correspondingly constructed transmission system . the invention has been described in detail with particular reference to preferred embodiments thereof and examples , but it will be understood that variations and modifications can be effected within the spirit and scope of the invention covered by the claims which may include the phrase “ at least one of a , b and c ” as an alternative expression that means one or more of a , b and c may be used , contrary to the holding in superquide v . directv , 69 uspq2d 1865 ( fed . cir . 2004 ). broadband radio access networks ( bran ); hiperlan type 2 , physical layer , etsi technical specification 101 475 v1 . 1 . 1 ( 2000 - 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