Patent Application: US-35695303-A

Abstract:
a communication apparatus and method that includes generating a series of successive spread - spectrum signals formed through the concatenation of a direct sequence spread spectrum code with a doppler tolerant polyphase code is herein disclosed . by transmitting concatenated codes , it is possible to demodulate received sequences that are very long compared to the inverse of the doppler frequency and / or the local oscillator drift . the method and apparatus may be applied to implement a wireless telecommunication system that efficiently transfers small quantities of data , such as status / telemetry , identification information and / or establishes device location at a low data rate over long distance in challenging radio environments and with low power consumption .

Description:
a critical problem in the above - described wireless communication scenario is the design of signature signals to be transmitted from a “ tag ” device and the detection algorithm that will permit fast acquisition and synchronization . although the present discussion related to “ tag ” devices , such as rfid / rtls ( real - time location systems ) tags or transponders , one of ordinary skill in the art will recognize that other compact , low power telemetry , identification , location - aware computing , data sensor networks , or location devices are also usability in this context . accordingly , the present invention is not limited to a particular type of tag or device , but rather is applicable to any device used in all such applications . in order to meet these challenges , one needs to maximize the radio link gain for the operational scenario , as , for example , when one party ( e . g ., a base station ) calls the other ( e . g ., an rfid tag ) and is to be uniquely identified by its id code . the parties need to establish a connection expeditiously , to exchange several hundreds or thousands of bits of information , measure distance , etc . and then get back to the power saving standby - receive mode . to meet this goal , communication can be performed at low bit rates ; for example , 50 – 500 bits per second ( bps ) can be a reasonable data rate . it is generally impractical to transmit low rate information in the same format as higher rate data because of the well - known effects of multipath fading and interference . multipath fading is the effect of the combining of multiple reflections which result in a reduction of the received signal at some frequencies and enhancement at others . interference is also a frequency selective effect . to mitigate the effects of a noisy environment , a low bandwidth signal typically employs a spread spectrum communications technique , one of a class of well - known techniques in which , although the data rate is low , the transmitted waveform is spread over a much wider bandwidth . such a system takes advantage of those frequencies that end up enhanced by multipath fading and interference in the channel . this is the case in many conventional military and commercial systems ; one commonly known system is the global positioning system ( gps ), where satellites transmit data at a bit rate of r b = 50 bps to terrestrial receivers while the signal is spread over a bandwidth of b = 1 mhz ( using the commercial , c / a code , system ). in the case of gps , the processing gain g p is approximately 43 db . a common issue with receiving high processing gain signals is that it can only be done with coherent detection ; coherent detection is often the only method of extracting weak signals from high levels of noise and interference . thus , in order to coherently demodulate the low rate data , the transmitter and receiver need to have very closely matched local oscillator ( lo ) frequencies , with the maximum acceptable difference of under ½ of the symbol rate . furthermore , because of the generally poor performance of commercially available los and the signal &# 39 ; s doppler component , a coherent receiver has to perform complex frequency search algorithms in order to synchronize with the transmitted waveform . for example , a gps receiver typically requires one to several minutes for the first acquisition and synchronization from a cold start . the faster receivers employ a fair amount of parallel processing , which is more costly and power hungry . however , if the gps receiver has location and time information , it can acquire faster , as it can estimate the satellite ephemerides , speed , and doppler frequency for all of the gps satellites it sees . generally , in a system transmitting 50 symbols / sec , the receiver needs to have its lo reference at less than 25 hz from the transmitter lo . but even a high quality local oscillator in a commercial receiver has a stability of approximately 1 part per million ( ppm ), or 900 hz at 900 mhz , making fast acquisition extremely difficult . as a particular example of this problem , when the frequency drift ( or deviation ) in the local lo , referred to herein as delta - f , is much larger than the symbol rate r s , a conventional receiver detector cannot find the data signal unless it performs a long and power consuming search . thus the traditional direct sequence spread spectrum ( dsss ) is impracticable for low power , low data rate , short messaging applications . a solution to these problems may be found in an unexpected area , namely the domain of radar technologies that deal with detecting signals with high doppler frequency uncertainty in a noisy environment . though long used for ranging applications , the applicability of radar - like waveforms at low power and low data rate communications has been largely overlooked to date . conventional radars transmit long pulses , up to fractions of a second in length , and then detect the reflections coherently and in real time even though they are shifted in frequency due to the doppler effect . as previously known , these long pluses are generated by very high power transmitters , often in the megawatt class . furthermore , radar pulses typically contain no communication data ; i . e ., the pulse does not convey information to another party or serve to exchange information ( other than by its very existence , of course ). thus , prior art high power radar pulse trains are not generally considered “ communications ” systems . in order to reach very far , radar has to transmit high energy pulses . as the peak transmission power is limited , the radar has to transmit long pulses to build - up the pulse energy , usually in the range of milliseconds . and , since the radar has to measure range and / or timing , the detector output has to be one very short pulse with energy equivalent to the entire pulse . the pulse width is inversely proportional to the signal bandwidth , i . e ., high timing accuracy translates into high signal bandwidth . so the radar problem is how to create long pulses — in the range of milliseconds — with signals that are spread in frequency over one or more mhz , and still be tolerant to doppler shift . this is actually the same problem in systems of interest here , where the main issue is that of identification , detection , and estimation of the timing and frequency shift . to address this problem , radars typically employ pulse compression codes to carry the pulse identification information the radar receiver needs to complete its ranging and detection function . pulse compression codes are digitally generated waveforms made of a long series of signal phase values . these signal phase values are also known as chips in the spread spectrum art . as the phases can take any value , they are called polyphase codes , as opposed to binary ones . these waveforms are generally of constant amplitude . when a such a long waveform is received and passed through a matched filter , the output of the filter is a short , high energy pulse representing the target , hence the name “ pulse compression .” information - bearing waveforms that can be coherently demodulated in the presence of a large doppler frequency shift belong to a class of waveforms called doppler tolerant polyphase codes ( dtpc ) or doppler tolerant pulse compression codes , which are derivatives of chirp waveforms . common examples of such codes in the literature are the lewis - kretschmer p4 ( see , for example , [ 15 ], [ 9 ], [ 10 ], [ 11 ]) and the nonlinear chirp p ( n , k ) ( see , for example , [ 12 ]). fig1 illustrates the principle of prior art pulse compression codes . the matched filter output clutter peaks 110 and 120 before and after the high pulse 130 are called sidelobes . the goal of a good system design is to create a high pulse - to - sidelobe ratio under a wide range of operating conditions and implementation tolerances . the doppler tolerant property resides in the fact that the pulse at the matched filter output and the pulse - to - sidelobe ratio does not deteriorate significantly when the received signal is shifted in frequency . the frequency shift translates in a time shift of the output pulse , a decrease of the pulse amplitude and an increase of the sidelobes . these effects are proportional with the ratio between the frequency shift and the pulse bandwidth . coming back to the gps example , a system based on dtpc would synchronize straight - on after the reception of several pulses , with no preliminary knowledge of the frequency difference . fig2 shows a simplified functional diagram of a system 200 according to the invention . in transmitter 202 , the data generating application 210 generates configuration information and data that is mapped 220 into position and / or phase for the sequence generator 225 . the signature control 223 controls the operating mode of the sequence generator 225 . this includes the types of codes used , the selection of identification codes in the two sections of the concatenated sequence , and switching between generating a dtpc concatenated sequence and a simple dsss sequence . the baseband signal is modulated onto a radio frequency ( rf ) carrier by means of the up - converter 230 , amplified by the power amplifier ( pa ) 235 and transmitted through the 240 antenna . the agile lo 232 provides the reference frequency to the up - converter 232 . in some embodiments , the system will have the capability to change its carrier frequency , thus the lo is shown as agile . receiver 203 includes a low noise amplifier ( lna ) 250 and the down - converter 252 that brings the signal into baseband . similarly to the transmitter side , the agile lo 252 provides the reference frequency for the downconverter . the matched filter 254 has to mirror the transmitted sequence . for this purpose , it is configured by the signature matching block 275 , which is controlled by the application 260 . in general the operation of the receiver can be described , but not limited to , two processes : detection / acquisition and demodulation . in the detection / acquisition process , the receiver attempts to detect a train of dtpc sequences . the configuration of the receiver matched filter 254 matches the codes transmitted by the sequence generator 225 and its output presents short pulses that expectedly rise above the noise . based on pulse position and / or phase information , the demodulator 256 acquires the timing and frequency of the received signal . these parameters enable the demodulation of data that is modulated on a transmitted dsss signal . the two processes are intertwined . the transmitter 202 , through its sequence generator 225 , transmits a combination of dtpc and dsss signals multiplexed either in time or in code domain , through the use of orthogonal codes ( see , for example , [ 13 ]). the id or signature info is superimposed on the sequence generator and the matched filter . in an exemplary embodiment , one device may include both the receiver 203 and transmitter 202 functions . furthermore , the matched filter 254 may be implemented in an application - specific integrated circuit ( asic ) of a type well - known the art . to solve the lo frequency drift problem , especially in situations where the transmitting lo is not particularly stable , the receiver uses a two - chirp differential calculation to resolve the frequency drift uncertainty . this is accomplished by configuring the transmitter to send a combination of conventional up - chirping pulses and down - chirping pulses . each chirp produces a pulse at the output of the receiver &# 39 ; s matched filter , separated in time by t o + d , where t o is the known spacing between the up - chirp and down - chirp signal , and delta - f is proportional to d . thus , using the two types of chirp waveforms as a “ pilot signal ,” the receiver can determine delta - f directly and thus synchronize the local lo to the transmitter lo , using conventional tuning devices or processes . once synchronized , the receiving device can use the delta - f information to properly modulate its own data transmissions , allowing rapid sync at the other end . due to distortions induced by the propagation channel and the rf amplification chain , the output of the matched filter is , in general , a cluster of correlation pulses and not a single pulse . therefore , an exemplary embodiment may involve combining the multitude of pulses into one equivalent pulse , at an equivalent timing . these techniques involve the use of , but are not limited to , rake receivers as known in the art of spread spectrum communications . the above - described fast synchronization aspect allows the present system to acquire signals even when the data symbol rate r s is much less than delta - f , i . e ., use of the two - chirp process both resolves the time - frequency ambiguity inherent in chirp waveforms and compensates for lo frequency uncertainty . as a further benefit , this method allows the receiver to determine chip sync from the mid - point of the upward - and downward - chirping waveforms , i . e ., the time between the two chirp &# 39 ; s correlation peaks . in an exemplary embodiment , when r s & lt ;& lt ; delta - f , the total synchronization time ( which is the total length of the pilot signal consisting in several dtpc sequences ) is still less than ( 15 / r s ). the matched filter complexity is roughly determined by the number of chips of the dtpc sequence , which is equal to the chip rate multiplied by the sequence duration . as the chip rate is roughly equal to the signal bandwidth , the bandwidth - duration product is referred as a measure of the matched filter or the sequence complexity . for example , a sequence with chip rate of 3 . 2768 mchip / s and duration of 20 ms would have a length l = 65 , 536 chips . a direct implementation would have to execute a complex multiply - add operation for each chip in the receiver matched filter at a rate equal to the chip rate , which is impracticable . an fft implementation for a filter of type p4 ( as described in , for example , [ 9 ], [ 10 ], and [ 11 ] but without output multipliers as described in [ 11 ]) would have to perform a number k of complex multiplications equal to : k = m 2 * log 2 ⁡ ( m ) + m - l ( 2 ) where l = m 2 . in our case , m = 256 = 2 8 , and the result is k = 1279 , which is still impracticable . thus there arises a need for sequence and matched filter configurations which have a high bandwidth - duration product , but involve a polyphase code of lower complexity . in an exemplary embodiment , a concatenation of two codes is used : a direct sequence code and a polyphase code . the sequence length is then the product of the two codes lengths . for example , a sequence of length 65 , 536 can be created by concatenating a direct sequence code of length 1024 with a polyphase similar to the dtpc one previously mentioned , but with a length of only 64 . thus m = 8 = 2 3 , and k is only 19 . the direct sequence matched filter is easier to implement as it involves only adders and sign changers . for the sake of clarity , the present disclosure refers to the overall sequence as concatenated dtpc . fig3 describes the timing of the concatenated codes , referred to herein as code 1 and code 2 . code 1 is the internal code , the one close to the rf channel . it repeats a direct sequence spreading code of n chips , which carries the id or signature data of interest . if the chip rate is r , the resulting repetition rate of direct sequence codewords is r / n . for example , for r = 3 . 2768 mchip / s and n = 1024 , the code 1 repetition rate would be 3 . 2 khz . the direct sequence matched filter can detect / demodulate coherently if the frequency drift is roughly less than ¼ of the repetition rate or about 800 hz in the present example . code 1 may be generated from a code family such as gold or kasami or a 4 - phase family ( see , for example , [ 13 ] [ 14 ]). code 1 increases the snr by approximately n times in samples of n chips periodicity . the chip phase resulting from the concatenation of two codes is shown in equation ( 3 ): as shown in fig3 , code 1 modulates consecutive chips 310 . it changes from chip to chip and repeats the same sequence of n values . code 2 is a dtpc polyphase code and it modulates consecutive code 1 codewords 320 . code 2 has a periodicity of n chips , and in an exemplary embodiment it can be a p4 code such as those described in [ 9 ], [ 10 ], or [ 11 ]. fig4 shows an exemplary implementation of a detector segment 400 using a matched filter for this waveform that can detect in one attempt . the matched filter 410 can operate when the frequency drift — the local oscillator frequency difference between the transmitter and receiver — is ¼ of the code 1 repetition rate or less . for higher values , a frequency search has to be performed . the detector 400 is composed of two matched filters . the internal one ( close to the rf / analog ) is matched to code 1 , and may be of a qpsk direct sequence type as ( in the general sense ) the direct sequence phase can take any of the values of 0 , 90 , 180 , or 270 degrees . the code 2 filter 410 performs the chirp - like or polyphase matched filtering over the whole sequence length of ( n · m )/ r seconds . both matched filters run at the sampled rate of r , but code 2 has an equivalent chip rate of r / n , therefore the taps from the shift register come every n positions . thresholds block 420 and decision block 430 analyze sequences at the rate of r samples per second to identify the peak patterns . in dispersive channels , threshold block 420 has to perform well - known rake receiver functions in order to minimize the effects of multipath . the order in which the steps of the present method are performed is purely illustrative in nature . in , fact , the steps can be performed in any order or in parallel , unless otherwise indicated by the present disclosure . the method of the present invention may be performed in either hardware , software , or any combination thereof , as those terms are currently known in the art . in particular , the present method may be carried out by software , firmware , or microcode operating on a computer or computers of any type . additionally , software embodying the present invention may comprise computer instructions in any form ( e . g ., source code , object code , interpreted code , etc .) stored in any computer - readable medium ( e . g ., rom , ram , magnetic media , punched tape or card , compact disc ( cd ) in any form , dvd , etc .). furthermore , such software may also be in the form of a computer data signal embodied in a carrier wave , such as that found within the well - known web pages transferred among devices connected to the internet . accordingly , the present invention is not limited to any particular platform , unless specifically stated otherwise in the present disclosure . while particular embodiments of the present invention have been shown and described , it will be apparent to those skilled in the art that changes and modifications may be made without departing from this invention in its broader aspect and , therefore , the appended claims are to encompass within their scope all such changes and modifications as fall within the true spirit of this invention . the following publications , cited by “[ number ]” above , are hereby incorporated herein by reference in their entireties . j . b . andersen , et al ., propagation measurements and models for wireless communications channels , ieee communications magazine , vol . 33 , iss . 1 , january 1995 , pp . 42 – 49 [ 2 ] goldsmith , a . j ., et al ., a measurement - based model for predicting coverage areas of urban microcells , ieee journal of selected areas in communications , vol . 11 , iss . 7 , september 1993 , pp . 1013 – 1023 [ 3 ] sklar , b ., rayleigh fading channels in mobile digital communication systems , ieee communications magazine , vol . 35 , iss . 7 , july 1997 , pp . 90 – 100 [ 4 ] tanis ii , w . j ., et al ., building penetration characteristics of 880 mhz and 1922 mhz radio waves , 43rd ieee vehicular technology conference , 1993 , pp . 206 – 209 [ 5 ] dres , d ., et al ., building penetration measurements for 2 . 4 ghz broadcasting cdma system , vehicular technology conference , 1999 , vol . 4 , pp . 1982 – 1987 [ 6 ] hendrickson , c ., et al ., wideband wireless peer to peer propagation measurements , record of the thirty - third asilomar conference on signals , systems , and computers , 1999 , vol . 1 , pp . 183 – 189 [ 7 ] kozono , s ., et al ., mobile propagation loss and delay spread characteristics with a low base station antenna on an urban road , ieee transactions on vehicular technology , february 1993 , vol . 42 , iss . 1 , pp . 103 – 109 [ 8 ] mogensen , p . e ., et al ., urban area radio propagation measurements at 955 and 1845 mhz for small and micro cells , global telecommunications conference , 1991 ( globecom &# 39 ; 91 ), pp . 1297 – 1302 [ 9 ] lewis , b . l ., et al ., a new class of polyphase pulse compression codes and techniques , ieee trans . on aerospace and elect . sys ., may 1981 , vol . aes - 17 , no . 3 [ 10 ] kretschmer , f . f ., et al ., sidelobe reduction techniques for polyphase pulse compression codes , record of the ieee 2000 international radar conference , pp . 416 – 421 [ 11 ] popovic , b . m ., efficient matched filter for the generalized chirp - like polyphase sequences , ieee transactions on aerospace and electronic systems , july 1994 , vol . 30 , iss . 3 , pp . 769 – 777 [ 12 ] felhauer , t ., design and analysis of new p ( n , k ) polyphase pulse compression codes , ieee transactions on aerospace and electronic systems , july 1994 , vol . 30 iss . 3 , pp . 865 – 874 [ 13 ] dinan , e . h ., et al ., spreading codes for direct sequence cdma and wideband cdma cellular networks , ieee communications magazine , september 1998 , vol . 36 , iss . 9 , pp . 48 – 54 [ 14 ] macdonald , t . g ., et al ., comparison of direct - sequence spread - spectrum multiple - access systems with qpsk data modulation , 1999 ieee military communications conference proceedings , vol . 1 , pp . 561 – 565 [ 15 ] kretschmer , f . f ., et al ., doppler properties of polyphase coded pulse compression waveform , ieee transactions on aerospace and electronic systems , vol . aes - 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