Patent Document

CROSS-REFERENCE TO RELATED APPLICATIONS 
     The present application is based upon and claims the benefit of priority from the prior Japanese Patent Application No. 2006-269630 filed on Sep. 29, 2006, with the Japanese Patent Office, the entire contents of which are incorporated herein by reference. 
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     It is related to semiconductor integrated circuits, and particularly relates to a voltage detecting circuit for detecting the level of a potential that needs to be monitored. 
     2. Description of the Related Art 
     Semiconductor integrated circuits such as DRAM may employ a boosted-voltage generating circuit for internally generating a boosted power supply potential and/or a negative-voltage generating circuit for generating a negative potential based on a power supply voltage having predetermined potentials supplied from an external source. In the case of DRAM, for example, a boosted-voltage generating circuit for supplying a boosted potential as a word-line activating potential higher than the HIGH level is used to reliably store the HIGH level in memory cells at high speed. Further, a negative voltage generated by a negative-voltage generating circuit is used in the memory cell array. 
       FIG. 1  is a drawing showing an example of the configuration of a DRAM including a boosted-voltage generating circuit and negative-voltage generating circuit. A DRAM  10  shown in  FIG. 1  includes an internal power supply generating unit  11 , a memory cell array  12 , and a memory access control circuit  13 . The memory access control circuit  13  performs a read and write operation with respect to the memory cell array  12  in response to control signals and address signals supplied from an external source. The internal power supply generating unit  11  includes a voltage detecting circuit  21  and a voltage generating circuit  22 . The voltage generating circuit  22  includes a boosted-voltage generating circuit  23  and a negative-voltage generating circuit  24 . A boosted potential and negative potential generated by the boosted-voltage generating circuit  23  and negative-voltage generating circuit  24  are supplied to the memory cell array  12 . The memory cell array  12  includes a transistor  25 , a capacitor  26 , a word-line driver  27 , a word line WL, and a bit line BL. A plurality of word lines and a plurality of bit lines are arranged in rows and columns. A transistor  25  and capacitor  26  connected to each intersecting point constitutes a memory cell for storing one bit. 
     The voltage detecting circuit  21  detects a boosted potential and negative potential supplied to the memory cell array  12 . Specifically, the voltage detecting circuit  21  compares a potential generated by dividing the boosted potential with a reference potential, and drives the boosted-voltage generating circuit  23  to step up the output of the boosted-voltage generating circuit  23  upon detecting that the divided potential drops below the reference potential. Further, the voltage detecting circuit  21  compares a potential generated by dividing the negative potential with a reference potential, and drives the negative-voltage generating circuit  24  to lower the output of the negative-voltage generating circuit  24  upon detecting that the divided potential rises above the reference potential. 
       FIG. 2  is a drawing showing an example of the circuit configuration of the voltage detecting circuit  21 . The circuit configuration shown in  FIG. 2  corresponds to the portion of the voltage detecting circuit  21  that relates to the detection of a boosted potential. 
     The voltage detecting circuit  21  of  FIG. 2  includes resistor elements R 1  and R 2 , a high-frequency-compensation capacitive element C 1 , and a differential amplifier  31 . The differential amplifier  31  operates as a comparator circuit for comparing two inputs, and has an output thereof supplied as a drive signal (activation signal) to the boosted-voltage generating circuit  23  so as to control the active/inactive state of the boosted-voltage generating circuit  23 . The inverted input node of the differential amplifier  31  receives a potential obtained by the resistor elements R 1  and R 2  dividing a boosted potential output from the boosted-voltage generating circuit  23 , and the non-inverted input of the differential amplifier  31  receives a reference potential generated by a reference potential generating circuit  29 . 
     As the output potential of the boosted-voltage generating circuit  23  drops due to current consumption in the memory cell array  12 , the above-noted divided potential becomes lower than the reference potential. In response to the divided potential lower than the reference potential, the differential amplifier  31  asserts the drive signal, which is its output signal. In response to the assertion of the drive signal, the boosted-voltage generating circuit  23  becomes active, thereby raising its output potential. As the divided potential becomes higher than the reference potential due to the rise of the output potential, the operation of the boosted-voltage generating circuit  23  comes to a halt. 
     In order to suppress needless current consumption, resistor elements having extremely large resistances are used as the resistor elements R 1  and R 2 . The amount of an electric current that actually flows is around 1 microampere. The divided potential appearing at the joint point between the resistor elements R 1  and R 2  thus does not respond with sufficient speed to a change in the boosted potential. There is thus a problem in that the response characteristics at high frequencies are not satisfactory. The high-frequency-compensation capacitive element C 1  is provided for the purpose of compensating for the response characteristics at high frequencies. The high-frequency-compensation capacitive element C 1  provides a low-impedance coupling between the boosted potential and the divided potential at high frequencies, thereby achieving a configuration in which a high-frequency fluctuation in the boosted potential directly propagates to the divided potential. This attains satisfactory response characteristics at high frequencies. 
       FIG. 3  is a drawing showing an example of the circuit configuration of the voltage detecting circuit  21 . The circuit configuration shown in  FIG. 3  corresponds to the portion of the voltage detecting circuit  21  that relates to the detection of a negative potential. 
     The voltage detecting circuit  21  of  FIG. 3  includes resistor elements R 3  and R 4 , a high-frequency-compensation capacitive element C 2 , and a differential amplifier  32 . The output of the differential amplifier  32  is supplied as a drive signal (activation signal) to the negative-voltage generating circuit  24  so as to control the active/inactive state of the negative-voltage generating circuit  24 . The non-inverted input node of the differential amplifier  32  receives a potential obtained by the resistor elements R 3  and R 4  dividing a negative potential output from the negative-voltage generating circuit  24 , and the inverted input of the differential amplifier  32  receives a reference potential generated by the reference potential generating circuit  29 . 
     The operation of the voltage detecting circuit  21  shown in  FIG. 3  is basically the same as that of the voltage detecting circuit  21  shown in  FIG. 2 . Like the high-frequency-compensation capacitive element C 1  shown in  FIG. 2 , the high-frequency-compensation capacitive element C 2  is provided for the purpose of compensating for response characteristics at high frequencies. The high-frequency-compensation capacitive element C 2  provides a low-impedance coupling between the negative potential and the divided potential at high frequencies, thereby achieving a configuration in which a high-frequency fluctuation in the negative potential directly propagates to the divided potential. This attains satisfactory response characteristics at high frequencies. 
     In the configurations shown in  FIG. 2  and  FIG. 3 , the resistor elements R 1  through R 4  may be implemented by use of a metal material or polysilicon material disposed in a metal layer or polysilicon layer. In recent years, however, the metal layer and polysilicon layer serving as signal interconnect layers have become low resistance for the purpose of increasing signal speed. Because of this, it is difficult to manufacture resistor elements having high resistance. If the resistor elements R 1  through R 4  are made by using N-type or P-type diffusion layers having high resistance, elements of relatively small size having desired resistance may be obtained. 
     The capacitive elements C 1  and C 2  may be implemented by placing an oxide film between a diffusion layer and a polysilicon layer or metal layer situated above the diffusion layer. Alternatively, a capacitor may be implemented by using an N-channel or P-channel MOS transistor. 
       FIGS. 4A and 4B  are drawings showing an example of a capacitive element implemented by using a P-channel MOS transistor.  FIG. 4A  is a plan view of the capacitive element, and  FIG. 4B  is a cross-sectional view of the capacitive element. 
     The configuration shown in  FIGS. 4A and 4B  includes metal interconnects  41  disposed in a metal layer for connection to a generated power supply, a polysilicon gate  42 , a gate contact  43 , source-drain contacts  44 , a P-type diffusion layer  45 , an N-type substrate  46 , and a metal interconnect  47  disposed in the metal layer for connection to a divided potential node. The metal interconnects  41  are coupled to the P-type diffusion layer  45  via the source-drain contacts  44 , and the metal interconnect  47  is coupled to the polysilicon gate  42  via the gate contact  43 . An oxide film is placed between the P-type diffusion layer  45  and the polysilicon gate  42 , thereby forming a capacitor between the P-type diffusion layer  45  and the polysilicon gate  42 . 
     A voltage detecting circuit having a similar configuration to that of the voltage detecting circuit  21  shown in  FIG. 2  is used in various circuits.  FIG. 5  is a drawing showing an example of the configuration of a DC-DC converter using a voltage detecting circuit. 
     A DC-DC converter  50  shown in  FIG. 5  includes a voltage detecting circuit  51 , switching-element-control circuits  52  and  53 , transistors  54  and  55 , an inductor  56 , and a capacitor  57 . The voltage detecting circuit  51  detects an output voltage VOut of the DC-DC converter  50 . Specifically, the voltage detecting circuit  51  compares a potential generated by dividing the output potential with a reference potential, and makes the transistor  54  conductive to raise the output potential upon detecting that the divided potential drops below the reference potential. Further, the voltage detecting circuit  51  makes the transistor  55  conductive to lower the output potential upon detecting that the divided potential rises above the reference potential. The voltage detecting circuit  51  operating as described above may be implemented by using substantially the same circuit configuration as that of the voltage detecting circuit  21  shown in  FIG. 21 . 
     In a voltage detecting circuit used in a semiconductor integrated circuit as described above, a capacitive element may be implemented by placing an oxide film between a diffusion layer and a polysilicon layer or metal layer, or may be implemented by use of an N-channel or P-channel MOS transistor. Either configuration, however, has a problem in that the circuitry size of a capacitive element is so large as to hinder the effort of reducing circuit size. 
     Accordingly, there is a need for a voltage detecting circuit that does not use a capacitive element in a semiconductor integrated circuit. 
     [Patent Document 1] Japanese Patent Application Publication No. 5-63147 
     [Patent Document 2] Japanese Patent Application Publication No. 2001-237374 
     SUMMARY OF THE INVENTION 
     It is a general object to provide a semiconductor integrated circuit that substantially obviates one or more problems caused by the limitations and disadvantages of the related art. 
     Features and advantages will be presented in the description which follows, and in part will become apparent from the description and the accompanying drawings, or may be learned by practice of the invention according to the teachings provided in the description. Objects as well as other features and advantages of the present invention will be realized and attained by a semiconductor integrated circuit particularly pointed out in the specification in such full, clear, concise, and exact terms as to enable a person having ordinary skill in the art to practice the invention. 
     To achieve these and other advantages in accordance with the purpose, the invention provides a semiconductor integrated circuit which includes a semiconductor substrate, one or more wells formed in the semiconductor substrate, one or more diffusion layers formed in the one or more wells, a plurality of interconnects formed in an interconnect layer, the one or more diffusion layers and the plurality of interconnects being connected in series to provide a coupling between a first potential and a second potential, and a comparison circuit coupled to one of the interconnects set at a third potential between the first potential and the second potential, and configured to compare the third potential with a reference potential, wherein a first interconnect of the plurality of interconnects that is set to the first potential is connected to at least a first well of the one or more wells and connected to a first diffusion layer of the one or more diffusion layers that is formed in the first well. 
     According to at least one embodiment, one or more diffusion layers constitute the resistor elements of a potential divider circuit, and one or more wells formed around the diffusion layers are coupled to a potential (i.e., the above-noted first potential) that is to be detected. This configuration makes it possible for parasitic capacitances between the one or more diffusion layers and the one or more wells to serve as a high-frequency-compensation parasitic capacitance, thereby providing a voltage detecting circuit that has a high-frequency-compensation capacitance without using a capacitive element. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Other objects and further features of the present invention will be apparent from the following detailed description when read in conjunction with the accompanying drawings, in which: 
         FIG. 1  is a drawing showing an example of the configuration of a DRAM including a boosted-voltage generating circuit and negative-voltage generating circuit; 
         FIG. 2  is a drawing showing an example of the circuit configuration of the voltage detecting circuit; 
         FIG. 3  is a drawing showing an example of the circuit configuration of the voltage detecting circuit; 
         FIGS. 4A and 4B  are drawings showing an example of a capacitive element implemented by using a P-channel MOS transistor; 
         FIG. 5  is a drawing showing an example of the configuration of a DC-DC converter using a voltage detecting circuit; 
         FIG. 6  is a drawing showing an example of the configuration of a first embodiment of a voltage detecting circuit according to the present invention; 
         FIGS. 7A and 7B  are drawings showing an example of the configuration of resistor elements and high-frequency-compensation parasitic capacitance; 
         FIG. 8  is a drawing showing an example of the configuration of a second embodiment of a voltage detecting circuit according to the present invention; 
         FIGS. 9A and 9B  are drawings showing an example of the configuration of resistor elements and high-frequency-compensation parasitic capacitance; 
         FIG. 10  is a voltage waveform diagram for explaining a voltage fluctuation in the case of the first embodiment; 
         FIG. 11  is a drawing showing an example of the configuration of a third embodiment of a voltage detecting circuit according to the present invention; 
         FIGS. 12A and 12B  are drawings showing an example of the configuration of resistor elements and parasitic capacitance; 
         FIG. 13  is a drawing showing an example of the configuration of a fourth embodiment of a voltage detecting circuit according to the present invention; 
         FIGS. 14A and 14B  are drawings showing an example of the configuration of resistor elements and parasitic capacitance; 
         FIG. 15  is a voltage waveform diagram for explaining a voltage fluctuation in the case of the third embodiment; 
         FIG. 16  is a voltage waveform diagram for explaining a voltage fluctuation in the case of the fourth embodiment; 
         FIG. 17  is a drawing showing an example of the configuration of a fifth embodiment of a voltage detecting circuit according to the present invention; 
         FIGS. 18A and 18B  are drawings showing an example of the configuration of resistor elements and parasitic capacitance; 
         FIGS. 19A and 19B  are drawings showing an example of the configuration of resistor elements and parasitic capacitance; 
         FIG. 20  is a drawing for explaining the effect of selective coupling of parasitic capacitances; 
         FIG. 21  is a drawing for explaining the effect of selective coupling of parasitic capacitances; and 
         FIG. 22  is a voltage waveform diagram for explaining a voltage fluctuation in the case of FIG.  21 -( c ). 
     
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     In the following, embodiments of the present invention will be described in detail with reference to the accompanying drawings. 
       FIG. 6  is a drawing showing an example of the configuration of a first embodiment of a voltage detecting circuit according to the present invention. For the sake of explanation, a configuration relevant to the detection of a boosted potential by the voltage detecting circuit is illustrated. 
     The voltage detecting circuit  60  of  FIG. 6  includes resistor elements R 1  and R 2 , a differential amplifier  61 , and a high-frequency-compensation parasitic capacitance  62 . The differential amplifier  61  operates as a comparator circuit for comparing two inputs, and has an output thereof supplied as a drive signal (activation signal) to the boosted-voltage generating circuit  23  so as to control the active/inactive state of the boosted-voltage generating circuit  23 . The inverted input node of the differential amplifier  61  receives a potential obtained by the resistor elements R 1  and R 2  dividing a boosted potential output from the boosted-voltage generating circuit  23 , and the non-inverted input of the differential amplifier  61  receives a reference potential generated by a reference potential generating circuit  29 . 
     As the output potential of the boosted-voltage generating circuit  23  drops due to current consumption in a load circuit, the above-noted divided potential becomes lower than the reference potential. In response to the divided potential lower than the reference potential, the differential amplifier  61  asserts the drive signal, which is its output signal. In response to the assertion of the drive signal, the boosted-voltage generating circuit  23  becomes active, thereby raising its output potential. As the divided potential becomes higher than the reference potential due to the rise of the output potential, the operation of the boosted-voltage generating circuit  23  comes to a halt. 
     In order to suppress needless current consumption, resistor elements having extremely large resistances are used as the resistor elements R 1  and R 2 . The amount of an electric current that actually flows is around 1 microampere. The divided potential appearing at the joint point between the resistor elements R 1  and R 2  thus does not respond with sufficient speed to a change in the boosted potential. The high-frequency-compensation parasitic capacitance  62  is parasitic to the resistor elements R 1  and R 2  formed of a diffusion layer as will later be described, and serves to compensate for response characteristics at high frequencies. The high-frequency-compensation parasitic capacitance  62  provides a low-impedance coupling between the boosted potential and the divided potential at high frequencies, thereby achieving a configuration in which a high-frequency fluctuation in the boosted potential directly propagates to the divided potential. This attains satisfactory response characteristics at high frequencies. 
       FIGS. 7A and 7B  are drawings showing an example of the configuration of the resistor elements R 1  and R 2  and high-frequency-compensation parasitic capacitance.  62  of the voltage detecting circuit  60 .  FIG. 7A  is a plan view of the resistor elements, and  FIG. 7B  is a cross-sectional view of the resistor elements. 
     The configuration shown in  FIGS. 7A and 7B  includes a metal interconnect  71 , metal interconnect  72 , and metal interconnect  73  disposed in a metal layer, and includes contacts  74  through  76 , a P-type diffusion layer  77 , a P-type substrate  78 , an N well  79 , an N region  80 , and a contact  81 . The N well  79  is formed in the P-type substrate  78 , and the P-type diffusion layer  77  is formed in the N well  79 . The metal interconnects  71  through  73  are connected to the P-type diffusion layer  77  via the respective contacts  74  through  76 . 
     The N region  80  for the purpose of potential coupling is provided in the N well  79 , and is connected to the metal interconnect  71  via the contact  81 . With this configuration, the potential of the N well  79  is set to the potential of the metal interconnect  71 . 
     The metal interconnect  71  is coupled to a boosted potential generated by the boosted-voltage generating circuit  23 , and corresponds to the position of a node A shown in  FIG. 6 . The metal interconnect  72  is coupled to a divided potential appearing at the joint point between the resistor elements R 1  and R 2 , and corresponds to the position of a node B shown in  FIG. 6 . The metal interconnect  73  is coupled to one end of the resistor element R 2  on the ground potential side, and corresponds to the position of a node C shown in  FIG. 6 . A single P-type diffusion layer  77  constitutes the resistor elements R 1  and R 2 , and the contact  75  situated halfway through the longitudinal extension of the P-type diffusion layer  77  serves to provide the divided potential. Namely, the contact  75  is connected to the P-type diffusion layer  77  at a position between the contact  74  of the metal interconnect  71  and the contact  76  of the metal interconnect  73 . The ratio of potential division can be adjusted by controlling the position of the contact  75 . 
     A parasitic capacitance exists between the P-type diffusion layer  77  and the N well  79 , and serves as the high-frequency-compensation parasitic capacitance  62 . The N well  79  is coupled to the boosted potential via the N region  80  as previously described, so that high-frequency-compensation parasitic capacitance  62  is coupled between the boosted potential and the P-type diffusion layer  77 . Namely, the high-frequency-compensation parasitic capacitance  62  is situated between the boosted potential and the resistor elements R 1  and R 2  as illustrated in  FIG. 6 . In  FIG. 6 , the high-frequency-compensation parasitic capacitance  62  is illustrated as if it was comprised of two separate capacitances for the sake of illustration. In reality, however, there is only a single capacitance covering over the entirety of the resistor elements R 1  and R 2  (i.e., the entirety of the P-type diffusion layer  77 ) inclusive of the junction point for the contact  75  (i.e., the node for detecting the divided potential) as shown in  FIG. 7 . 
     In the first embodiment of the voltage detecting circuit according to the present invention, the P-type diffusion layer constitutes the resistor elements of the potential divider circuit, and the N well formed around the P-type diffusion layer is coupled to the potential that is to be detected (i.e., the boosted potential in this example). This configuration makes it possible for the parasitic capacitance between the P-type diffusion layer and the N well to serve as a high-frequency-compensation parasitic capacitance, thereby providing a voltage detecting circuit that has a high-frequency-compensation capacitance without using a capacitive element. 
       FIG. 8  is a drawing showing an example of the configuration of a second embodiment of a voltage detecting circuit according to the present invention.  FIG. 8  shows a configuration relevant to the detection of a negative potential by the voltage detecting circuit. 
     The voltage detecting circuit  85  of  FIG. 8  includes resistor elements R 3  and R 4 , a differential amplifier  86 , and a high-frequency-compensation parasitic capacitance  87 . The differential amplifier  86  operates as a comparator circuit for comparing two inputs, and has an output thereof supplied as a drive signal (activation signal) to the negative-voltage generating circuit  24  so as to control the active/inactive state of the negative-voltage generating circuit  24 . The non-inverted input node of the differential amplifier  86  receives a potential obtained by the resistor elements R 3  and R 4  dividing a negative potential output from the negative-voltage generating circuit  24 , and the inverted input of the differential amplifier  86  receives a reference potential generated by the reference potential generating circuit  29 . The operation of the voltage detecting circuit  85  shown in  FIG. 8  is basically the same as that of the voltage detecting circuit  60  shown in  FIG. 6 . 
       FIGS. 9A and 9B  are drawings showing an example of the configuration of the resistor elements R 3  and R 3  and high-frequency-compensation parasitic capacitance  87  of the voltage detecting circuit  85 .  FIG. 8A  is a plan view of the resistor elements, and  FIG. 8B  is a cross-sectional view of the resistor elements. 
     The configuration shown in  FIGS. 9A and 9B  includes a metal interconnect  91 , metal interconnect  92 , and metal interconnect  93  disposed in a metal layer, and includes contacts  94  through  96 , an N-type diffusion layer  97 , a P-type substrate  98 , an N well  99 , a P well  100 , a P region  101 , a contact  102 , and a contact  103 . The N well  99  is formed in the P-type substrate  98 , with the P well  100  formed in the N well  99 , and the N-type diffusion layer  97  formed in the P well  100 . The metal interconnects  91  through  93  are connected to the N-type diffusion layer  97  via the respective contacts  94  through  96 . 
     The P region  101  for the purpose of potential coupling is provided in the P well  100 , and is connected to the metal interconnect  91  via the contact  102 . With this configuration, the potential of the P well  100  is set to the potential of the metal interconnect  91 . 
     The metal interconnect  91  is coupled to a negative potential generated by the negative-voltage generating circuit  24 , and corresponds to the position of a node D shown in  FIG. 8 . The metal interconnect  92  is coupled to a divided potential appearing at the joint point between the resistor elements R 3  and R 4 , and corresponds to the position of a node E shown in  FIG. 8 . The metal interconnect  93  is coupled to one end of the resistor element R 3  on the power supply potential side, and corresponds to the position of a node F shown in  FIG. 8 . A single N-type diffusion layer  97  constitutes the resistor elements R 3  and R 4 , and the contact  95  situated halfway through the longitudinal extension of the N-type diffusion layer  97  serves to provide the divided potential. The ratio of potential division can be adjusted by controlling the position of the contact  95 . 
     A parasitic capacitance exists between the N-type diffusion layer  97  and the P well  100 , and serves as the high-frequency-compensation parasitic capacitance  87 . The P well  100  is coupled to the negative potential via the P region  101  as previously described, so that high-frequency-compensation parasitic capacitance  87  is coupled between the negative potential and the N-type diffusion layer  97 . Namely, the high-frequency-compensation parasitic capacitance  87  is situated between the negative potential and the resistor elements R 3  and R 4  as illustrated in  FIG. 8 . In  FIG. 8 , the high-frequency-compensation parasitic capacitance  87  is illustrated as if it was comprised of two separate capacitances for the sake of illustration. In reality, however, there is only a single capacitance covering over the entirety of the resistor elements R 3  and R 4  (i.e., the entirety of the N-type diffusion layer  97 ) inclusive of the junction point for the contact  95  (i.e., the node for detecting the divided potential) as shown in  FIG. 9 . 
     In the second embodiment of the voltage detecting circuit according to the present invention, the N-type diffusion layer constitutes the resistor elements of the potential divider circuit, and the P well formed around the N-type diffusion layer is coupled to the potential that is to be detected (i.e., the negative potential in this example). This configuration makes it possible for the parasitic capacitance between the N-type diffusion layer and the P well to serve as a high-frequency-compensation parasitic capacitance, thereby providing a voltage detecting circuit that has a high-frequency-compensation capacitance without using a capacitive element. 
     In the first or second embodiment described above, the parasitic capacitance in existence between the P-type diffusion layer  77  and the N well  79  or between the N-type diffusion layer  97  and the P well  100  serves as a high-frequency-compensation parasitic capacitance. If the capacitance value of such parasitic capacitance is too large, low impedance appears even with respect to low-frequency changes in the detected potential. In such a case, the divided potential changes to follow a low-frequency change in the detected potential, so that the divided potential may be set to a different potential than a true divided potential that should be produced by the resistor-based potential division. 
       FIG. 10  is a voltage waveform diagram for explaining a voltage fluctuation in the case of the first embodiment. In  FIG. 10 , a voltage waveform  110  illustrates a fluctuation in the boosted potential, and a voltage waveform  111  illustrates a fluctuation in the divided potential. 
     As the boosted potential drops due to current consumption by the driving of a load circuit as shown by the voltage waveform  110 , the divided potential also drops as shown by the voltage waveform  111  due to capacitive coupling through the high-frequency-compensation parasitic capacitance  62 . Thereafter, the amount of electric charge in the high-frequency-compensation parasitic capacitance  62  gradually changes, so that the divided potential changes until it reaches the potential defined by the ratio of the resistor elements R 1  and R 2 . Such a change is illustrated as a waveform portion  112 . Upon the start of operation of the boosted-voltage generating circuit  23  after a predetermined response time of the differential amplifier circuit, the boosted potential rises as shown by the voltage waveform  110 , so that the divided potential having a capacitive coupling to the boosted potential rises substantially in the same manner as the boosted potential. As the divided potential exceeds the reference potential, the operation of the boosted-voltage generating circuit  23  comes to a halt after a predetermined response time of the differential amplifier circuit. Thereafter, the amount of electric charge in the high-frequency-compensation parasitic capacitance  62  gradually changes, so that the divided potential changes until it reaches the potential defined by the ratio of the resistor elements R 1  and R 2 . Such a change is illustrated as a waveform portion  113 . 
     In this manner, the capacitive coupling through the high-frequency-compensation parasitic capacitance  62  causes the divided potential to change in such a manner as to follow a change in the boosted potential, thereby improving the response characteristics of the voltage detecting circuit in terms of boosted-potential fluctuation. If the capacitance value of the high-frequency-compensation parasitic capacitance  62  is too large, however, the divided potential ends up following low-frequency fluctuation of the boosted potential. Specifically, the speed at which the divided potential changes due to a change in the amount of electric charge stored in the high-frequency-compensation parasitic capacitance  62  becomes slower (i.e., the slope of the waveform portions  112  and  113  becomes gentler), so that a next-phase change in the boosted potential may occur before the divided potential reaches the target potential defined by the ratio of the resistor elements R 1  and R 2 . In this case, the next change in the boosted potential starts before the divided potential reaches the target potential, resulting in the accumulation of errors between the divided potential and the target potential. 
     Accordingly, it is desirable to provide a configuration in which the capacitance value of the high-frequency-compensation parasitic capacitance can be lowered in the configuration of the first and second embodiments. In the following, such configuration will be described. 
       FIG. 11  is a drawing showing an example of the configuration of a third embodiment of a voltage detecting circuit according to the present invention. In  FIG. 11 , the same elements as those of  FIG. 6  are referred to by the same numerals, and a description thereof will be omitted. 
     The voltage detecting circuit  60 A of  FIG. 11  includes resistor elements R 1  and R 2 , a differential amplifier  61 , a high-frequency-compensation parasitic capacitance  62 A, and a parasitic capacitance  62 B. The high-frequency-compensation parasitic capacitance  62 A is parasitic to the resistor element R 1  formed of a diffusion layer as will later be described, and serves to compensate for response characteristics at high frequencies. The parasitic capacitance  62 B is parasitic to the resistor element R 2  formed of a diffusion layer. 
       FIGS. 12A and 12B  are drawings showing an example of the configuration of the resistor elements R 1  and R 2  and parasitic capacitances  62 A and  62 B of the voltage detecting circuit  60 A.  FIG. 12A  is a plan view of the resistor elements, and  FIG. 12B  is a cross-sectional view of the resistor elements. In  FIGS. 12A and 12B , the same elements as those of  FIGS. 7A and 7B  are referred to by the same numerals, and a description thereof will be omitted. 
     The configuration shown in  FIGS. 12A and 12B  includes a metal interconnect  71 , metal interconnect  72 , and metal interconnect  73  disposed in a metal layer, and includes contacts  74 A and  75 B, contact  76 A and  76 B, a P-type diffusion layer  77 A, an N-type diffusion layer  77 B, a P-type substrate  78 , an N well  79 , an N region  80 , a contact  81 , a P well  82 , and a P region  83 . The N well  79  is formed in the P-type substrate  78 , and the P-type diffusion layer  77 A and the P-well  82  are formed in the N well  79 . Further, the N-type diffusion layer  77 B is formed in the P-well  82 . The metal interconnect  71  is connected to the P-type diffusion layer  77 A via the contact  74 . The metal interconnect  72  is connected to the P-type diffusion layer  77 A via the contact  75 A, and is also connected to the N-type diffusion layer  77 B via the contact  75 B. The metal interconnect  73  is connected to the N-type diffusion layer  77 B via the contact  76 A, and is also connected to the P region  83  of the P-well  82  via the contact  76 B. 
     The N region  80  for the purpose of potential coupling is provided in the N well  79 , and is connected to the metal interconnect  71  via the contact  81 . With this configuration, the potential of the N well  79  is set to the potential of the metal interconnect  71 . Further, the metal interconnect  73  is connected to the P region  83  of the P-well  82  via the contact  76 B as described above, so that the potential of the P-well  82  is set to the potential of the metal interconnect  73 . 
     A parasitic capacitance exists between the P-type diffusion layer  77 A and the N well  79 , and serves as the high-frequency-compensation parasitic capacitance  62 A. The N well  79  is coupled to the boosted potential via the N region  80  as previously described, so that high-frequency-compensation parasitic capacitance  62 A is coupled between the boosted potential and the P-type diffusion layer  77 A. Namely, the high-frequency-compensation parasitic capacitance  62 A is situated between the boosted potential and the resistor element R 1  (and the node for detecting a divided potential) as illustrated in  FIG. 11 . 
     Further, a parasitic capacitance exists between the N-type diffusion layer  77 B and the P-well  82 . The P-well  82  is coupled to the ground potential via the P region  83 , so that this parasitic capacitance is coupled between the ground potential and the N-type diffusion layer  77 B. Namely, the parasitic capacitance  62 B is situated between the ground potential and the resistor element R 2  (and the node for detecting a divided potential) as illustrated in  FIG. 11 . 
     In the third embodiment of the voltage detecting circuit according to the present invention, the P-type diffusion layer and N-type diffusion layer constitutes the resistor elements of the potential divider circuit, and the N well formed around the P-type diffusion layer is coupled to the potential that is to be detected (i.e., the boosted potential in this example), with the P well formed around the N-type diffusion layer being coupled to a fixed potential (i.e., the ground potential in this example). This configuration makes it possible for the parasitic capacitance between the P-type diffusion layer and the N well to serve as a high-frequency-compensation parasitic capacitance, and also couples the parasitic capacitance of the resistor element (i.e., N-type diffusion layer) on the ground side to the ground potential, thereby preventing the capacitance of the high-frequency-compensation capacitance from becoming larger than necessary. 
       FIG. 13  is a drawing showing an example of the configuration of a fourth embodiment of a voltage detecting circuit according to the present invention. In  FIG. 13 , the same elements as those of  FIG. 8  are referred to by the same numerals, and a description thereof will be omitted. 
     The voltage detecting circuit  85 A of  FIG. 13  includes resistor elements R 3  and R 4 , a differential amplifier  86 , a high-frequency-compensation parasitic capacitance  87 A, and a parasitic capacitance  87 B. The high-frequency-compensation parasitic capacitance  87 A is parasitic to the resistor element R 4  formed of a diffusion layer as will later be described, and serves to compensate for response characteristics at high frequencies. The parasitic capacitance  87 B is parasitic to the resistor element R 3  formed of a diffusion layer. 
       FIGS. 14A and 14B  are drawings showing an example of the configuration of the resistor elements R 3  and R 4  and parasitic capacitances  87 A and  87 B of the voltage detecting circuit  85 A.  FIG. 14A  is a plan view of the resistor elements, and  FIG. 14B  is a cross-sectional view of the resistor elements. 
     The configuration shown in  FIGS. 14A and 14B  includes a metal interconnect  91 , metal interconnect  92 , and metal interconnect  93  disposed in a metal layer, and includes a contact  94 ,  95 A,  95 B, and  96 , an N-type diffusion layer  97 A, a P-type diffusion layer  97 B, a P-type substrate  98 , an N well  99 , a P well  100 , a P region  101 , a contact  102 , a contact  103 , and an N region  104 . The N well  99  is formed in the P-type substrate  98 , and the P-type diffusion layer  97 B and the P well  100  are formed in the N well  99 , with the N-type diffusion layer  97 A formed in the P well  100 . 
     The metal interconnect  91  is connected to the N-type diffusion layer  97 A via the contact  94 , and is also connected to the P region  101  of the P-well  100  via the contact  102 . The metal interconnect  92  is connected to the N-type diffusion layer  97 A via the contact  95 A, and is also connected to the P-type diffusion layer  97 B via the contact  95 B. The metal interconnect  93  is connected to the P-type diffusion layer  97 B via the contact  96 , and is also connected to the N region  104  of the N-well  99  via the contact  103 . 
     With these connecting, the potential of the P well  100  is set to the potential of the metal interconnect  91 . Further, the potential of the N well  99  is set to the potential of the metal interconnect  93 . The metal interconnect  91  is coupled to a negative potential generated by the negative-voltage generating circuit  24 , and the metal interconnect  93  is coupled to one end of the resistor element R 3  on the power supply potential side. 
     A parasitic capacitance exists between the N-type diffusion layer  97 A and the P well  100 , and serves as the high-frequency-compensation parasitic capacitance  87 A. The P well  100  is coupled to the negative potential via the P region  101  as previously described, so that high-frequency-compensation parasitic capacitance  87 A is coupled between the negative potential and the N-type diffusion layer  97 A. Namely, the high-frequency-compensation parasitic capacitance  87 A is situated between the negative potential and the resistor element R 4  (and the node for detecting a divided potential) as illustrated in  FIG. 13 . 
     Further, a parasitic capacitance exists between the P-type diffusion layer  97 B and the N-well  99 . The N-well  99  is coupled to the power supply potential via the N region  104 , so that this parasitic capacitance is coupled between the power supply potential and the P-type diffusion layer  97 B. Namely, the high-frequency-compensation parasitic capacitance  87 B is situated between the power supply potential and the resistor element R 3  (and the node for detecting a divided potential) as illustrated in  FIG. 13 . 
     In the fourth embodiment of the voltage detecting circuit according to the present invention, the N-type diffusion layer and P-type diffusion layer constitutes the resistor elements of the potential divider circuit, and the P well formed around the N-type diffusion layer is coupled to the potential that is to be detected (i.e., the negative potential in this example), with the N well formed around the P-type diffusion layer being coupled to a fixed potential (i.e., the power supply potential in this example). This configuration makes it possible for the parasitic capacitance between the N-type diffusion layer and the P well to serve as a high-frequency-compensation parasitic capacitance, and also couples the parasitic capacitance of the resistor element (i.e., P-type diffusion layer) on the power supply potential side to the power supply potential, thereby preventing the capacitance of the high-frequency-compensation capacitance from becoming larger than necessary. 
       FIG. 15  is a voltage waveform diagram for explaining a voltage fluctuation in the case of the third embodiment. In  FIG. 15 , a voltage waveform  120  illustrates a fluctuation in the boosted potential, and a voltage waveform  121  illustrates a fluctuation in the divided potential. 
     In the third embodiment, the capacitance value of the high-frequency-compensation parasitic capacitance is set to an appropriate value as previously described, so that the divided potential is set to a potential defined by the ratio of resistor-based potential division as far as low-frequency fluctuation is concerned, which is comparable to the operation cycles of the boosted-voltage generating circuit and the voltage detecting circuit. Namely, although the divided potential changes due to capacitive coupling between the boosted potential node and the divided potential node, capacitive impedance relevant to the low-frequency fluctuation comparable to the circuit operation cycles becomes larger than the resistance of the resistor elements constituting the potential divider circuit. This means that the divided potential is properly set to a potential defined by the ratio of resistor-based potential division. 
       FIG. 16  is a voltage waveform diagram for explaining a voltage fluctuation in the case of the fourth embodiment. In  FIG. 16 , a voltage waveform  122  illustrates a fluctuation in the negative potential, and a voltage waveform  123  illustrates a fluctuation in the divided potential. In this case, the polarity of waveforms is reversed due to the replacement of a boosted potential with a negative potential. Except for the reversal, the voltage fluctuation waveforms are substantially the same as those shown in  FIG. 15 . Namely, as in the third embodiment, the high-frequency-compensation parasitic capacitance is set to an appropriate value in the fourth embodiment, so that the divided potential is properly set to a potential defined by the ratio of resistor-based potential division. 
     In the configurations of the first through fourth embodiments described above, the capacitance value of the high-frequency-compensation parasitic capacitance is fixedly set at the time of circuit design. In order to properly set the capacitance value of high-frequency-compensation parasitic capacitance, it is desirable that the capacitance value is adjustable according to need. In the following, such configuration will be described. 
       FIG. 17  is a drawing showing an example of the configuration of a fifth embodiment of a voltage detecting circuit according to the present invention. In  FIG. 17 , the same elements as those of  FIG. 6  and  FIG. 8  are referred to by the same numerals, and a description thereof will be omitted. 
     A voltage detecting circuit  60 B of  FIG. 17  includes resistor elements R 1 - 1 , R 1 - 2 , R 2 - 1 , and R 2 - 2 , a differential amplifier  61 , and parasitic capacitances  62 - 1  through  62 - 4 . The parasitic capacitances  62 - 1  through  62 - 4  are parasitic to the resistor elements R 1 - 1 , R 1 - 2 , R 2 - 1 , and R 2 - 2 , respectively, which are formed of diffusion layers as will later be described. A voltage detecting circuit  85 B of  FIG. 17  includes resistor elements R 3 - 1 , R 3 - 2 , R 4 - 1 , and R 4 - 2 , a differential amplifier  86 , and parasitic capacitances  87 - 1  through  87 - 4 . The parasitic capacitances  87 - 1  through  87 - 4  are parasitic to the resistor elements R 4 - 2 , R 4 - 1 , R 3 - 2 , and R 3 - 1 , respectively, which are formed of diffusion layers as will later be described. 
     In  FIG. 17 , connections illustrated by dotted lines indicate circuit portions for which connection or disconnection is selectable. Provision is thus made such that the size of capacitance serving as high-frequency-compensation parasitic capacitance is adjustable by selecting whether to connect individual parasitic capacitances to the boosted power supply side or to the resistor-series side. A choice between connection and disconnection may be made by mask switching, or may be made by selectively cutting one of the connections through laser exposure. In so doing, the portion disconnected by the laser beam may be implemented as a fuse. 
       FIGS. 18A and 18B  are drawings showing an example of the configuration of the resistor elements and parasitic capacitances of the voltage detecting circuit  60 B.  FIG. 18A  is a plan view of the resistor elements, and  FIG. 18B  is a cross-sectional view of the resistor elements. 
     The configuration shown in  FIGS. 18A and 18B  includes metal interconnects  130  through  135  disposed in a metal layer, P-type diffusion layers  136  through  139 , a metal interconnect  140 , connection/disconnection-selectable interconnects  141  through  143 , a P-type substrate  150 , N wells  151  through  154 , and N regions  155  through  158 . The N wells  151  through  154  are formed in the P-type substrate  150 , and the P-type diffusion layers  136  through  139  are formed in the N wells  151  through  154 , respectively. The metal interconnects  131  through  135  are connected to the P-type diffusion layers  136  through  139  via contacts. Through such connections, the P-type diffusion layers  136  through  139  are connected in series via the intervening metal interconnects  132  through  134 . 
     The metal interconnects  140  through  143  are respectively connected to the N regions  155  through  158  of the N wells  151  through  154  via contacts. The connection/disconnection-selectable interconnects  141  through  143  are electrically connected through selective cutting to either the metal interconnect  130  or the metal interconnects  132  through  134 . For the sake of convenience of explanation, the connection/disconnection-selectable interconnect  143  is illustrated as being connected to both the metal interconnect  130  and the metal interconnect  134 . Either a fuse portion  143   a  or fuse portion  143   b  of the connection/disconnection-selectable interconnect  143  is cut by a laser beam, for example, thereby performing selective coupling. 
     The metal interconnect  130  is coupled to a boosted potential generated by the boosted-voltage generating circuit  23 . The P-type diffusion layers  136  through  139  correspond to the resistor elements R 1 - 1 , R 1 - 2 , R 2 - 1 , and R 2 - 2 , respectively, shown in  FIG. 17 . 
     Parasitic capacitances exist between the P-type diffusion layers  136  through  139  and the N wells  151  through  154 , and are shown as parasitic capacitances  62 - 1  through  62 - 4 , respectively, in  FIG. 17 . Selection of a connection state of the connection/disconnection-selectable interconnects  141  through  143  makes it possible to selectively couple the parasitic capacitances  62 - 2  through  62 - 4  to either the boosted potential or the resister series. Those of the parasitic capacitances  62 - 2  through  62 - 4  electrically coupled to the boosted potential serve as a high-frequency-compensation parasitic capacitance together with the parasitic capacitance  62 - 1 . Strictly speaking, those of the parasitic capacitances  62 - 2  through  62 - 4  electrically coupled to the resistor series also provide capacitive couplings. Since such couplings are connected to potentials having dropped from the boosted potential through the resistor series, however, their function as a high-frequency-compensation parasitic capacitance is relatively small. 
       FIGS. 19A and 19B  are drawings showing an example of the configuration of the resistor elements and parasitic capacitances of the voltage detecting circuit  85 B shown in  FIG. 17 .  FIG. 19A  is a plan view of the resistor elements, and  FIG. 19B  is a cross-sectional view of the resistor elements. 
     The configuration shown in  FIGS. 19A and 19B  includes metal interconnects  160  through  165  disposed in a metal layer, N-type diffusion layers  166  through  169 , a metal interconnect  170 , connection/disconnection-selectable interconnects  171  through  173 , a P-type substrate  180 , P wells  181  through  184 , P regions  185  through  188 , and an N well  189 . The P wells  181  through  184  are formed in the N well  189  of the P-type substrate  180 , and the N-type diffusion layers  166  through  169  are formed in the P wells  181  through  184 , respectively. The metal interconnects  161  through  165  are connected to the N-type diffusion layers  166  through  169  via contacts. Through such connections, the N-type diffusion layers  166  through  169  are connected in series via the intervening metal interconnects  162  through  164 . 
     The metal interconnects  170  through  173  are respectively connected to the P regions  185  through  188  of the P wells  181  through  184  via contacts. The connection/disconnection-selectable interconnects  171  through  173  are electrically connected through selective cutting to either the metal interconnect  160  or the metal interconnects  162  through  164 . 
     The metal interconnect  160  is coupled to a negative potential generated by the negative-voltage generating circuit  24 . The N-type diffusion layers  166  through  169  correspond to the resistor elements R 4 - 2 , R 4 - 1 , R 3 - 2 , and R 3 - 1 , respectively, shown in  FIG. 17 . 
     Parasitic capacitances exist between the N-type diffusion layers  166  through  169  and the P wells  181  through  184 , and are shown as parasitic capacitances  87 - 1  through  87 - 4 , respectively, in  FIG. 17 . Selection of a connection state of the connection/disconnection-selectable interconnects  171  through  173  makes it possible to selectively couple the parasitic capacitances  87 - 2  through  87 - 4  to either the negative potential or the resister series. Those of the parasitic capacitances  87 - 2  through  87 - 4  electrically coupled to the negative potential serve as a high-frequency-compensation parasitic capacitance together with the parasitic capacitance  87 - 1 . Strictly speaking, those of the parasitic capacitances  87 - 2  through  87 - 4  electrically coupled to the resistor series also provide capacitive couplings. Since such couplings are connected to potentials having dropped from the boosted potential through the resistor series, however, their function as a high-frequency-compensation parasitic capacitance is relatively small. 
       FIG. 20  and  FIG. 21  are drawings for explaining the effect of selective coupling of parasitic capacitances. In  FIG. 20  and  FIG. 21 , the same elements as those of  FIG. 17  are referred to by the same numerals, and a description thereof will be omitted. 
     When all the parasitic capacitances  62 - 1  through  62 - 4  are electrically coupled to the boosted potential as shown in FIG.  20 -( a ), such circuit corresponds to an equivalent circuit shown in FIG.  20 -( b ). All the resistor elements R 1 - 1 , R 1 - 2 , R 2 - 1 , and R 2 - 2  have the same resistance R, and all the parasitic capacitances  62 - 1  through  62 - 4  have the same capacitance value C, for example. The circuit shown in FIG.  20 -( b ) is equivalent to the circuit illustrated in FIG.  20 -( c ). The circuit shown in FIG.  20 -( c ) is substantially the same as the circuit shown in  FIG. 6 , and, thus, produces the same voltage fluctuation waveforms as shown in  FIG. 10 . 
     When all the parasitic capacitances  62 - 2  through  62 - 4  are electrically coupled to the resistor series as shown in FIG.  21 -( a ), such circuit corresponds to an equivalent circuit shown in FIG.  21 -( b ). All the resistor elements R 1 - 1 , R 1 - 2 , R 2 - 1 , and R 2 - 2  have the same resistance R, and all the parasitic capacitances  62 - 1  through  62 - 4  have the same capacitance value C, for example. The circuit shown in FIG.  21 -( b ) is equivalent to the circuit illustrated in FIG.  21 -( c ). The circuit shown in FIG.  21 -( c ) is substantially the same as the circuit shown in  FIG. 11 , and the capacitance value is a small value, i.e., C/2. Since the capacitance is small in this case, voltage waveforms as shown in  FIG. 22  may be obtained. 
       FIG. 22  is a voltage waveform diagram for explaining a voltage fluctuation in the case of FIG.  21 -( c ). In  FIG. 22 , a voltage waveform  190  illustrates a fluctuation in the boosted potential, and a voltage waveform  191  illustrates a fluctuation in the divided potential. 
     Since capacitive coupling between the boosted potential node and the divided potential node is small, the divided potential drops only slightly as shown by the voltage waveform  191  despite that fact that the boosted potential shown as the voltage waveform  190  has a sudden drop due to current consumption by a load circuit. Thereafter, the divided voltage gradually drops due to an electric current flowing through the diffused resistors having large resistance, and is set to a potential defined by the ratio of resistor-based potential division. 
     Further, the present invention is not limited to these embodiments, but various variations and modifications may be made without departing from the scope of the present invention.

Technology Category: 5