Patent Document

BACKGROUND OF THE INVENTION  
       [0001]     1. Field of the Invention  
         [0002]     The present invention relates to a receiving device and an integrated circuit for reception.  
         [0003]     2. Description of the Related Art  
         [0004]     Digital audio radio services in the U.S. are called “DARS”, and in DARS, satellite waves and terrestrial waves are used in combination so that even a receiver mounted in a mobile unit such as vehicle can reliably receive the radio waves.  
         [0005]     More specifically, in the DARS, a 2.3 GHz band is used, and as shown in part B of  FIG. 6 , two services are broadcast. Currently, each of the services uses a frequency band of 12.5 MHz. As is also shown in part A of  FIG. 6 , one service is formed of two ensembles A and B, and each of these ensembles A and B provides 50 channels of programs contents. Therefore, one service provides programs of 100 channels.  
         [0006]     The ensemble A is broadcast with individual signals A 1 , A 2 , and A 3 , and the ensemble B is broadcast with individual signals B 1 , B 2 , and B 3 . That is, the contents of the signals A 1 , A 2 , and A 3  are the same, and the contents of the signals B 1 , B 2 , and B 3  are the same. Therefore, if any one of the signals A 1 , A 2 , and A 3  can be received, the program of the ensemble A can be listened to, and in a similar manner, if any one of the signals B 1 , B 2 , and B 3  can be received, the program of the ensemble B can be listened to.  
         [0007]     As is also shown in part A of  FIG. 6 , the signals A 1  to A 3  and B 1  to B 3  are arranged as the signals A 1 , A 2 , A 3 , B 3 , B 2 , and B 1  in order of frequency, and the signals A 1 , A 2 , and A 3 , and the signals B 3 , B 2 , and B 1  are symmetrically placed about a center frequency fC between the signal A 3  and the signal B 3 .  
         [0008]     The signals A 1 , A 2 , B 1 , and B 2  are QPSK (Quadrature Phase Shift Keying) signals. The signals A 1  and B 1  are transmitted from a broadcasting satellite BS 1  over the Western U.S., and the signals A 2  and B 2  are transmitted from a broadcasting satellite BS 2  over the Eastern U.S. (strictly speaking, the satellites BS 1  and BS 2  are positioned along the Equator at longitudes corresponding to the Western U.S. and the Eastern U.S.). Also, the signals A 3  and B 3  are OFDM (Orthogonal Frequency Division Multiplex) signals and are transmitted from an antenna on the ground.  
         [0009]     Therefore, since the signals A 1 , A 2 , B 1 , and B 2  are satellite waves, and a diversity effect can be obtained by the satellites BS 1  and BS 2 , a broadcast can be listened to over the entire U.S. Also, when there is a high-rise building, radio waves are sometimes blocked, but this is compensated for by the signals A 3  and B 3  of the terrestrial waves. Therefore, even when the receiving conditions of radio waves of a receiver mounted in a vehicle greatly change as the vehicle travels, it is possible to satisfactorily receive a broadcast.  
         [0010]     In DARS, since the signals A 1  to A 3  and B 1  to B 3  are broadcast by frequency division in the above-described manner, a receiver therefor is constructed as shown in, for example,  FIG. 7 . In the following description, for brevity of explanation, as shown in  FIG. 8A , the signals A 1  and A 2  are collectively denoted as A 12 , and the signals B 1  and B 2  are collectively denoted as B 12 .  
         [0011]     More specifically, in  FIG. 7 , the signals A 12 , A 3 , B 12 , and B 3  are received by an antenna  11 , and the received signals A 12  to B 3  are supplied to a first mixer circuit  14  via a band-pass filter  12  and a high-frequency amplifier  13 . Furthermore, a first local oscillation signal SLO is supplied from a first local oscillation circuit  15  to the first mixer circuit  14 , whereby the signals A 12  to B 3  are frequency-converted into first intermediate frequency signals.  
         [0012]     When the ensemble A is to be listened to (when the signals A 1  to A 3  are subjects to be received), as indicated by the solid line in  FIG. 8A , the first local oscillation signal SLO is set to a predetermined frequency fL which is lower than those of the signals A 12  and A 3 . Therefore. as shown in  FIG. 8B , the signal A 12  is frequency-converted into a first intermediate frequency signal SIF 12  (at intermediate frequency fIF 12 ), the signal A 3  is frequency-converted into a first intermediate frequency signal SIF 3  (at intermediate frequency fIF 3 ), and the signals B 12  and B 3  are frequency-converted into first intermediate frequency signals SIF 45  and SIF 6 , respectively.  
         [0013]     When the image rejection characteristics are taken into consideration, the first intermediate frequencies fIF 12  and fIF 3  cannot be decreased too much, and since a frequency band of 2.3 GHz is used in a broadcast, the first intermediate frequencies fIF 12  and fIF 3  are set to 100 MHz or higher. For example, the following are set:  
         [0014]     fIF 12  is approximately 113 MHz, and fIF 3  is approximately 116 MHz  
         [0015]     Also, when the ensemble B is to be listened to (when the signals B 1  to B 3  are subjects to be received), as indicated by the broken line in  FIG. 8A , the first local oscillation signal SLO is set to a predetermined frequency fH which is higher than those of the signals B 12  and B 3 . Therefore, as shown in  FIG. 8C , the signal B 12  is frequency-converted into a first intermediate frequency signal SIF 12  (at intermediate frequency fIF 12 ), the signal B 3  is frequency-converted into a first intermediate frequency signal SIF 3  (at intermediate frequency fIF 3 ), and the signals A 12  and A 3  are frequency-converted into first intermediate frequency signals SIF 45  and SIF 6 , respectively.  
         [0016]     Therefore, when any one of the ensembles A and B is to be listened to, the intermediate frequency signals SIF 12  to SIF 6  are supplied to a band-pass filter  21 L for a first intermediate-frequency filter, whereby an intermediate frequency signal SIF 12  is extracted. Then, this signal is supplied to a second mixer circuit  22 L, a second local oscillation signal having a predetermined frequency is provided from a second local oscillation circuit  23 , and this signal is supplied to the mixer circuit  22 L, whereby the signal SIF 12  is frequency-converted into a second intermediate frequency signal. Then, this signal is supplied to a demodulation circuit  25 L via a variable gain amplifier  24 L for AGC (Automatic Gain Control), whereby a digital audio signal of the target program is demodulated, and this signal is supplied to a selecting/combining circuit  26 .  
         [0017]     Also, the signals SIF 12  to SIF 6  from the first mixer circuit  14  is supplied to a band-pass filter  21 H for a first intermediate frequency filter, whereby the intermediate frequency signal SIF 3  is extracted. Then, this signal is supplied to a second mixer circuit  22 H, and furthermore, a second local oscillation signal from the second local oscillation circuit  23  is supplied to the mixer circuit  22 H, whereby the signal SIF 3  is frequency-converted into a second intermediate frequency signal. Then, this signal is supplied to a demodulation circuit  25 H via a variable gain amplifier  24 H for AGC, whereby a digital audio signal of the target program is demodulated, and this signal is supplied to the-selecting/combining circuit  26 .  
         [0018]     Then, in the selecting/combining circuit  26 , the signal from the demodulation circuit  25 L and the signal from the demodulation circuit  25 H are selected or combined, and is output at an output terminal  27 .  
         [0019]     Therefore, as a result of switching the frequency of the first local oscillation signal SLO to a frequency fL or a frequency fH, a digital signal of the ensemble A or a digital signal of the ensemble B is output at the terminal  27 .  
         [0020]     Then, at that time, when the ensemble A is received, since the digital signal demodulated from the received signal A 12  and the digital signal demodulated from the received signal A 3  are selected or combined, and is taken out at the terminal  27 , a digital signal having a small amount of error can be obtained regardless of the receiving conditions. Furthermore, also when the ensemble B is received, a digital signal having a small amount of error can be obtained regardless of the receiving conditions for the same reasons.  
         [0021]     However, in the above-described receiver, when the ensemble is switched from the ensemble A to the ensemble B, it is necessary to change the frequency of the first local oscillation signal SLO from the frequency fL to the frequency fH. That is, as is also clear from  FIGS. 8A  to  8 C, it is necessary to change the frequency of the first local oscillation signal SLO to a frequency larger than the occupied bandwidth 12.5 MHz of the services of the signals A 1  to A 3  and B 1  to B 3 . Also, the same applies to a case in which the ensemble is changed from the ensemble B to the ensemble A.  
         [0022]     The amount of change of this frequency is equal to or more than 10% of the frequencies fL and fH. Moreover, when the first local oscillation circuit  15  is formed by a PLL (Phase-Locked Loop), it is necessary to allow for some margin with respect to the range of change of the oscillation frequency of the VCO (Voltage Controlled Oscillator) of the PLL. For this reason, it is necessary to increase the range of change of the oscillation frequency of the VCO by making the resonance device of the VCO changeable. As a result, the construction becomes complex, and the phase noise characteristics of the local oscillation signal SLO deteriorate, causing the error rate of the digital signal to become worse.  
         [0023]     Also, as long as the first local oscillation circuit  15  is formed by a PLL, it takes time to change the frequency, and the ensemble cannot be received during that change.  
         [0024]     In addition, the first intermediate frequencies fIF 12  and fIF 3  are increased to 100 MHz or higher in the above-described manner, and as shown in  FIGS. 8B and 8C , it is necessary for the filters  21 L and  21 H to extract the first intermediate frequency signals SIF 12  and SIF 3  from within the crowded signals. As a result, the filters  21 L and  21 H are formed by an SAW (Surface Acoustic Wave) filter. For this reason, the cost increases, and when the circuit is formed into an IC (integrated circuit), the SAW filter must be provided externally. Furthermore, this becomes an obstacle to the reduction in size of the receiver.  
         [0025]     Also, when the demodulation of the demodulation circuits  25 L and  25 H is to be performed by a digital process, an intermediate frequency signal supplied to the demodulation circuits  25 L and  25 H must be formed into a frequency at which a digital process is possible. For this purpose, as is also shown in  FIG. 7 , for the receiving method, a double conversion method must be used, the construction becomes complex, and the number of parts is increased.  
       SUMMARY OF THE INVENTION  
       [0026]     The present invention aims to solve such problems as those described above.  
         [0027]     Accordingly, an object of the present invention is to provide a receiving device comprising: a receiving circuit for receiving a first signal and a second signal which are transmitted at mutually different frequencies; a circuit for forming first and second local oscillation signals, whose frequencies are both the center frequency between the first signal and the second signal, and whose phases differ by 90° from each other; a first mixer circuit for frequency-converting the received signal received by the receiving circuit into a first intermediate frequency signal in accordance with the first local oscillation signal; a second mixer circuit for frequency-converting the received signal received by the receiving circuit into a second intermediate frequency signal in accordance with the second local oscillation signal; a first phase-shift circuit to which the first intermediate frequency signal is supplied; a second phase-shift circuit to which the second intermediate frequency signal is supplied, in which the amount of the phase shift differs by 90° from that of the first phase-shift circuit; and an addition/subtraction circuit for performing one of addition and subtraction between the output signal of the first phase-shift circuit and the output signal of the second phase-shift circuit, wherein, by switching the process in the addition/subtraction circuit to the addition or the subtraction, the intermediate frequency signal corresponding to the first signal or the intermediate frequency signal corresponding to the second signal is selectively extracted from the addition/subtraction circuit.  
         [0028]     Therefore, while the local oscillation frequency is being fixed, the first signal or the second signal is selected.  
         [0029]     In particular, a receiving device is provided which is suitable for a case in which each of the first and second signals is formed of a signal of a plurality of programs, and the signals of individual programs are transmission programs which are arranged according to frequency symmetrically with respect to the center frequency.  
         [0030]     More specifically, when the ensemble is to be switched, since the frequency of the local oscillation signal does not need to be changed, the local oscillation circuit does not become complex. Also, the deterioration of the phase noise characteristics of the local oscillation signal, and the deterioration of the error rate of the digital signal do not occur. Furthermore, when the ensemble is to be switched, the switching can be performed easily at high speed, and the problem where the ensemble cannot be received during the switching, like when the local oscillation frequency is to be changed, does not occur.  
         [0031]     Another object of the present invention is to provide a reception integrated circuit which is suitable for constructing the above-described receiving device. According to the integrated circuit of the present invention, in addition to the above-described features, the intermediate-frequency filter can be formed by an active filter, and can be integrally formed into a one-chip IC with other circuits. This is effective in reducing the cost and the size of the receiver. Furthermore, even when demodulation is to be performed by a digital process, a single conversion may be used for the receiving method, the construction becomes simple, and the number of parts is decreased.  
         [0032]     The above and further objects, aspects and novel features of the invention will become more fully apparent from the following detailed description when read in conjunction with the accompanying drawings. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0033]      FIG. 1  is a block diagram showing an embodiment of the present invention;  
         [0034]      FIGS. 2A, 2B , and  2 C are frequency spectrum diagrams illustrating the present invention;  
         [0035]      FIG. 3  is a block diagram showing another embodiment of the present invention;  
         [0036]      FIG. 4  is a circuit diagram showing a part of the other embodiment of the present invention;  
         [0037]      FIG. 5  is a circuit diagram showing a part of the other embodiment of the present invention;  
         [0038]      FIG. 6  is a frequency spectrum diagram illustrating DARS;  
         [0039]      FIG. 7  is a block diagram showing the present invention; and  
         [0040]      FIGS. 8A, 8B , and  8 C are frequency spectrum diagrams illustrating the circuit of  FIG. 7 . 
     
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0041]      FIG. 1  shows an example of a DARS receiving circuit according to the present invention, in which a portion  30  surrounded by a one-dot chain line is formed into a one-chip IC. Signals A 1  to A 3 , and B 1  to B 3  are received by an antenna  51 , and the received signals A 1  to B 3  are supplied to mixer circuits  32 I and  32 Q via a band-pass filter  52 , which is formed of, for example, an SAW filter and which has a passing bandwidth of 12.5 MHz and furthermore via a high-frequency amplifier  31 .  
         [0042]     In a local oscillation circuit  33 , as shown in  FIG. 2A , an oscillation signal SLO having a frequency equal to the center frequency fC between the signal A 3  and the signal B 3  is formed, this signal SLO is supplied to a phase processing circuit  34 , whereby two local oscillation signals SLI and SLQ, whose phases differ by 90° from each other, with the frequency being kept at the value fC, are formed, and these signals SLI and SLQ are supplied to the mixer circuits  32 I and  32 Q, respectively.  
         [0043]     In the following description, for brevity of explanation, it is assumed that, as shown in  FIG. 2A , the signal SA represents each of the signals A 1  to A 3 , and the signal SB represents each of the signals B 1  to B 3 . That is, it is assumed that SA=A 1 , SA=A 2 , or SA=A 3 , and that SB=B 1 , SB=B 2 , or SB=B 3 . Then, it is arranged that: 
 
 SA=EA ·sin ω At  
 
 SB=EB ·sin ω Bt  
 
 where EA is the amplitude of the signal SA, EB is the amplitude of the signal SB, ωA is the angular frequency of the signal SA, and ωB is the angular frequency of the signal SB. 
 
 Also, it is arranged that: 
 
 SLI=EL ·sin ω Ct  
 
 SLQ=EL ·cos ω Ct  
 
 where EL is the amplitude of the signals SLI and SLQ, and ωC=2πfC. 
 
         [0044]     Then, from the mixer circuits  32 I and  32 Q, signals SIFI and SIFQ as described below are extracted:  
       SIFI   =         (     SA   +   SB     )     ×   SLI     ⁢     
     ⁢           =           EA   ·   sin     ⁢           ⁢   ω   ⁢           ⁢   At   ×     EL   ·   sin     ⁢           ⁢   ω   ⁢           ⁢   Ct     +       EB   ·   sin     ⁢           ⁢   ω   ⁢           ⁢   Bt   ×     EL   ·   sin     ⁢           ⁢   ω   ⁢           ⁢   Ct       ⁢     
     ⁢           =       α   ⁢     {         cos   ⁡     (       ω   ⁢           ⁢   A     -     ω   ⁢           ⁢   C       )       ⁢   t     -       cos   ⁡     (       ω   ⁢           ⁢   A     +     ω   ⁢           ⁢   C       )       ⁢   t       }       +     β   ⁢     {         cos   ⁡     (       ω   ⁢           ⁢   B     -     ω   ⁢           ⁢   C       )       ⁢   t     -     
     ⁢           ⁢       cos   ⁡     (       ω   ⁢           ⁢   B     +     ω   ⁢           ⁢   C       )       ⁢   t       }                 
       SIFQ   =         (     SA   +   SB     )     ×   SLQ     ⁢     
     ⁢           =           EA   ·   sin     ⁢           ⁢   ω   ⁢           ⁢   At   ×     EL   ·   cos     ⁢           ⁢   ω   ⁢           ⁢   Ct     +       EB   ·   sin     ⁢           ⁢   ω   ⁢           ⁢   Bt   ×     EL   ·   cos     ⁢           ⁢   ω   ⁢           ⁢   Ct       ⁢     
     ⁢           =       α   ⁢     {         sin   ⁡     (       ω   ⁢           ⁢   A     +     ω   ⁢           ⁢   C       )       ⁢   t     +       sin   ⁡     (       ω   ⁢           ⁢   A     -     ω   ⁢           ⁢   C       )       ⁢   t       }       +     β   ⁢     {         sin   ⁡     (       ω   ⁢           ⁢   B     +     ω   ⁢           ⁢   C       )       ⁢   t     +     
     ⁢           ⁢       sin   ⁡     (       ω   ⁢           ⁢   B     -     ω   ⁢           ⁢   C       )       ⁢   t       }                 
 
 where α=EA·EL/2, and β=EB·EL/2 
 
         [0045]     As will be described later, of the signals SIFI and SIFQ, the signal components of angular frequencies (ωA−ωC) and (ωB−ωC) are used as the intermediate frequency signals, and the signal components of angular frequencies (ωA+ωC) and (ωB+ωC) are removed by the intermediate frequency filter. Therefore, for the sake of simplicity, if the signal components of angular frequencies (ωA+ωC) and (ωB+ωC) to be removed are ignored, the above equations become: 
 
 SIFI =α·cos(ω A−ωC ) t +β·cos(ω B−ωC ) t  
 
 SIFQ =α·sin(ω A−ωC ) t +β·sin(ω B−ωC ) t  
 
         [0046]     Here, if it is arranged that ωA=ωC−Δω with regard to the signal SA, since, as is also shown in  FIG. 2A , the signal SA and the signal SB are symmetrically distributed about the frequency fC, the following equation holds: 
 
 ωB=ωC+Δω 
 
         [0047]     Then, if these equations are substituted in the equations for the signals SIFI and SIFQ, the following equations are obtained:  
       SIFI   =           α   ·     cos   ⁡     (       ω   ⁢           ⁢   C     -     Δ   ⁢           ⁢   ω     -     ω   ⁢           ⁢   C       )         ⁢   t     +       β   ·     cos   ⁡     (       ω   ⁢           ⁢   C     +     Δ   ⁢           ⁢   ω     -     ω   ⁢           ⁢   C       )         ⁢   t       ⁢     
     ⁢           =           α   ·     cos   ⁡     (       -   Δ     ⁢           ⁢   ω     )         ⁢   t     +       β   ·   cos     ⁢           ⁢   Δ   ⁢           ⁢   ω   ⁢           ⁢   t       ⁢     
     ⁢           =         α   ·   cos     ⁢           ⁢   Δ   ⁢           ⁢   ω   ⁢           ⁢   t     +       β   ·   cos     ⁢           ⁢   Δ   ⁢           ⁢   ω   ⁢           ⁢   t               
       SIFQ   =           α   ·     sin   ⁡     (       ω   ⁢           ⁢   C     -     Δ   ⁢           ⁢   ω     -     ω   ⁢           ⁢   C       )         ⁢   t     +       β   ·     sin   ⁡     (       ω   ⁢           ⁢   C     +     Δ   ⁢           ⁢   ω     -     ω   ⁢           ⁢   C       )         ⁢   t       ⁢     
     ⁢           =           α   ·     sin   ⁡     (       -   Δ     ⁢           ⁢   ω     )         ⁢   t     +       β   ·   sin     ⁢           ⁢   Δ   ⁢           ⁢   ω   ⁢           ⁢   t       ⁢     
     ⁢           =           -   α     ·   sin     ⁢           ⁢   Δ   ⁢           ⁢   ω   ⁢           ⁢   t     +       β   ·   sin     ⁢           ⁢   Δ   ⁢           ⁢   ω   ⁢           ⁢   t               
 
         [0048]     These signals SIFI and SIFQ are then supplied to phase-shift circuits  35 I and  35 Q. The phase-shift circuits  35 I and  35 Q are formed by an active filter in which, for example, a capacitor, a resistor, and an operational amplifier are used. The phase-shift circuit  35 I phase-shifts the signal SIFI by a value φ (φ is an arbitrary value), and the phase-shift circuit  35 Q phase-shifts the signal SIFQ by a value (φ+90°).  
         [0049]     In this manner, the phase-shift circuits  35 I and  35 Q cause the signal SIFQ to lead the signal SIFI by 90°, and the following equations hold:  
       SIFI   =         α   ·   cos     ⁢           ⁢   Δ   ⁢           ⁢   ω   ⁢           ⁢   t     +       β   ·   cos     ⁢           ⁢   Δ   ⁢           ⁢   ω   ⁢           ⁢   t           
       SIFQ   =           -   α     ·     sin   ⁡     (       Δ   ⁢           ⁢   ω   ⁢           ⁢   t     +     90   ⁢   °       )         +     β   ·     sin   ⁡     (       Δ   ⁢           ⁢   ω   ⁢           ⁢   t     +     90   ⁢   °       )           ⁢     
     ⁢           =           -   α     ·   cos     ⁢           ⁢   Δ   ⁢           ⁢   ω   ⁢           ⁢   t     +       β   ·   cos     ⁢           ⁢   Δ   ⁢           ⁢   ω   ⁢           ⁢   t             
 
 Therefore, between the signal SIFI and the signal SIFQ, the signal components α·cos Δωt are at the opposite phase from each other, and the signal components β·cos Δωt are in phase. 
 
         [0050]     These signals SIFI and SIFQ are then supplied to an addition/subtraction circuit  36 , and a control signal SSW is supplied from a terminal  37  to the addition/subtraction circuit  36 . This control signal SSW controls the operation of the addition/subtraction circuit  36  in such a way that when the program of the ensemble A is to be listened to, the addition/subtraction circuit  36  acts as a subtraction circuit, and when the program of the ensemble B is to be listened to, the addition/subtraction circuit  36  acts as an addition circuit.  
         [0051]     Therefore, a signal SIF such as that described below is extracted from the addition/subtraction circuit  36  in such a manner as to correspond to the control signal SSW. That is, during subtraction, the following is extracted:  
         SIF   =       SIFI   -   SIFQ     ⁢     
     ⁢           =       2   ⁢     α   ·   cos     ⁢           ⁢   Δ   ⁢           ⁢   ω   ⁢           ⁢   t     ⁢     
     ⁢           =       EL   ·   EA   ·   cos     ⁢           ⁢   Δ   ⁢           ⁢   ω   ⁢           ⁢   t           ,       
 
 and during addition, the following is extracted:  
       SIF   =       SIFI   +   SIFQ     ⁢     
     ⁢           =       2   ⁢     β   ·   cos     ⁢           ⁢   Δ   ⁢           ⁢   ω   ⁢           ⁢   t     ⁢     
     ⁢           =       EL   ·   EB   ·   cos     ⁢           ⁢   Δ   ⁢           ⁢   ω   ⁢           ⁢   t             
 
         [0052]     Here, the signal SIF=EL·EA·cos Δωt which is obtained during subtraction is, as is also shown in  FIG. 2B , the same intermediate frequency signal when the signal SA is received. The signals SIF 1  to SIF 3  contained in this signal SIF are the intermediate frequency signals of the signals A 1  to A 3 . Also, the signal SIF=EL·EB cos Δωt which is obtained during addition is, as is also shown in  FIG. 2C , the same intermediate frequency signal when the signal SB is received. The signals SIF 1  to SIF 3  contained in this signal SIF are the intermediate frequency signals of the signals B 1  to B 3 .  
         [0053]     Therefore, this signal SIF is supplied to a band-pass filter  41 H for an intermediate-frequency filter having passing characteristics such as those indicated by the broken line in, for example,  FIGS. 2B and 2C , whereby an intermediate frequency signal SIF 3  of a terrestrial-wave signal A 3  or B 3  is extracted. At this time, the intermediate frequency signals SIF 1  and SIF 2  and the above-mentioned signal components of angular frequencies (ωA+ωC) and (ωB+ωC) are removed by the band-pass filter  41 H.  
         [0054]     Then, this intermediate frequency signal SIF 3  is supplied to a demodulation circuit  43 H via a variable gain amplifier  42 H for AGC, whereby a digital audio signal of the target program is demodulated, and this signal is supplied to a selecting/combining circuit  44 .  
         [0055]     Also, the signal SIF from the addition/subtraction circuit  36  is supplied to a band-pass filter  41 L for an intermediate-frequency filter having passing characteristics such as those indicated by the broken line in, for example,  FIGS. 2B and 2C , whereby intermediate frequency signals SIF 2  and SIF 1  of the satellite-wave signals A 1  and A 2 , or B 1  and B 2  are extracted. At this time, the intermediate frequency signal SIF 3  and the above-mentioned signal components of angular frequencies (ωA+ωC) and (ωB+ωC) are removed by the filter  41 L.  
         [0056]     Then, these intermediate frequency signals SIF 2  and SIF 1  are supplied to a demodulation circuit  43 L via a variable gain amplifier  42 L for AGC, whereby a digital audio signal of the target program is demodulated, and this signal is supplied to the selecting/combining circuit  44 .  
         [0057]     Then, in the selecting/combining circuit  44 , the digital signal from the demodulation circuit  43 H and the digital signal from the demodulation circuit  43 L are selected or combined according to the received status of the signals A 1  to B 3 , and is extracted at an output terminal  45 . Of course, when it is desired to give priority to a receiving environment of a mobile unit in which the receiver is mounted and to satellite-wave reception, the AGC voltage obtained from the level detection circuit  46 L may be supplied as a gain control signal.  
         [0058]     At this time, parts of the intermediate frequency signals from the demodulation circuits  43 H and  43 L are supplied to level detection circuits  46 H and  46 L, whereby AGC voltages are formed, and these AGC voltages are supplied, as gain control signals, to the amplifiers  42 H and  42 L, whereby AGC is performed.  
         [0059]     In addition, although the level variation of the satellite wave is relatively small, the level variation of the terrestrial wave is relatively large. Therefore, for the high-frequency amplifier  31 , a variable gain amplifier is used, and the AGC voltage obtained from the level detection circuit  46 H is supplied, as a gain control signal, to the amplifier  31 , whereby AGC is performed.  
         [0060]     In this manner, according to the receiving circuit of  FIG. 1 , a broadcast by DARS can be received, and in a case where the ensemble is switched between the ensemble A and the ensemble B, the frequency fC of the local oscillation signals SLI and SLQ does not need to be changed. Consequently, the local oscillation circuit  33  may be formed in a standard construction and does not become complex. Also, since the phase noise characteristics of the local oscillation signals SLI and SLQ are not decreased, the error rate of the digital signal does not become worse.  
         [0061]     In addition, when the ensemble is to be switched, the addition/subtraction circuit  36  need only be switched to an addition operation or a subtraction operation. Consequently, the switching can be performed at high speed, and the problem of not being able to receive the ensemble during switching time does not occur.  
         [0062]     As is also clear from  FIGS. 2B and 2C , since the upper-limit frequency of the occupied bandwidth of the intermediate frequency signal SIF is equal to a half of the bandwidth of one ensemble, and the center frequencies of the filters  41 H and  41 L become approximately 1.3 MHz and 4.4 MHz, it is possible to form each of the filters  41 H and  41 L by an active filter. Therefore, it is possible to form the entirety into a one-chip IC as an IC  30 , excluding a band-pass filter  52  at the antenna input stage, and this is effective in reducing the costs and the size of the receiver.  
         [0063]     In addition, since the intermediate frequency of the intermediate frequency signals SIF 3  to SIF 1  is as low as several MHz, even when the demodulation of the demodulation circuits  43 H and  43 L is performed by a digital process, as shown in, for example,  FIG. 1 , for the receiving method, a single conversion may be used, the construction becomes simple, and the number of parts is decreased.  
         [0064]     In the receiving circuit shown in  FIG. 3 , a case is shown in which, by inverting or non-inverting the phase of the local oscillation signal SLQ when the ensemble A is received and when the ensemble B is received, the signals SIFI and SIFQ are always added together.  
         [0065]     More specifically, in the receiving circuit in  FIG. 3 , the control signal SSW is supplied as a phase control signal to the phase processing circuit  34 , so that the phase of the local oscillation signal SLQ is controlled such that: 
 
 SLQ=+EL ·cos ω Ct  . . . when the ensemble B is received, 
 
 and 
 
 SLQ=−EL ·cos ω Ct  . . . when the ensemble A is received. 
 
 The phase of the local oscillation signal SLI is fixed, as described above: 
 
 SLI=EL·sin ωCt  
 
         [0066]     In place of the addition/subtraction circuit  36  in  FIG. 1 , an addition circuit  38  is provided, and the signals SIFI and SIFQ output from the phase-shift circuits  35 I and  35 Q are supplied to the addition circuit  38 .  
         [0067]     According to such a construction, in the case of SLQ=+EL·cos ωCt, in the addition circuit  38 , the signal SIFI and the signal SIFQ are added together. Therefore, as is described with reference to the receiving circuit of  FIG. 1 , the signal SIF extracted from the addition circuit  38  becomes as follows:  
       SIF   =       SIFI   +   SIFQ     ⁢     
     ⁢           =       EL   ·   EB   ·   cos     ⁢           ⁢   Δ   ⁢           ⁢   ω   ⁢           ⁢   t           
 
 Therefore, it is possible to listen to the program of the ensemble B. 
 
         [0068]     On the other hand, in the case of SLQ=−EL·cos ωCt, the output signal of the phase-shift circuit  35 Q becomes the signal −SIFQ. Therefore, since, in the addition circuit  38 , subtraction between the signal SIFI and the signal SIFQ is performed, as is described with reference to the receiving circuit of  FIG. 1 , the signal SIF extracted from the addition circuit  38  becomes:  
       SIF   =       SIFI   -   SIFQ     ⁢     
     ⁢           =       EL   ·   EA   ·   cos     ⁢           ⁢   Δ   ⁢           ⁢   ω   ⁢           ⁢   t           
 
 Therefore, it is possible to listen to the program of the ensemble A. 
 
         [0069]     In this way, also in the receiving circuit of  FIG. 3 , a DARS broadcast can be received. In particular, according to the receiving circuit of  FIG. 3 , in a case where the ensemble is switched between the ensemble A and the ensemble B, it is only necessary to invert or non-invert the phase of the local oscillation signal SLQ by the phase processing circuit  34 . Therefore, the ensemble can be switched quickly. Also, since the phase-shift circuits  35 I and  35 Q and the addition circuit  38  can be formed by a poly-phase filter, the phase characteristics of the signal SIFI and the signal ±SIFQ can be improved.  
         [0070]     In  FIG. 4 , a case is shown in which the phase of the intermediate frequency signal SIFI is constant regardless of the ensemble which is received, but the phase of the intermediate frequency signal SIFQ is inverted or non-inverted between when the ensemble A is to be received and when the ensemble B is to be received.  
         [0071]     More specifically, the mixer circuit  32 Q is formed as a double balanced-type by transistors Q 321  to Q 327 . The received signals A 1  to A 3  and B 1  to B 3  are extracted as a balanced type from the amplifier  31  and are supplied to transistors Q 322  and Q 323 . Furthermore, the local oscillation signal SLQ is extracted as a balanced type from the phase processing circuit  34  and is supplied to transistors Q 324 , Q 327 , Q 325 , and Q 326 .  
         [0072]     Consequently, the intermediate frequency signal SIFQ is extracted as a balanced type from the mixer circuit  32 Q. That is, for example, the intermediate frequency signal +SIFQ is extracted from the transistors Q 324  and Q 326 , and the intermediate frequency signal −SIFQ is extracted from the transistors Q 325  and Q 327 .  
         [0073]     Then, these intermediate frequency signal ±SIFQ are supplied to a switching circuit  39 . This switching circuit  39  is formed as a balanced type by transistors Q 391  to Q 397 , and the intermediate frequency signals ±SIFQ which are supplied thereto are supplied to a phase-shift circuit  36 Q in accordance with the control signal SSW with the phase kept as it is or with the phase being inverted.  
         [0074]     More specifically, based on the control signal SSW, when the transistor Q 395  is on and transistor Q 396  is off, the transistors Q 392  and Q 393  are turned on, and the transistors Q 391  and Q 394  are turned off. Therefore, the intermediate frequency signal +SIFQ extracted from the transistors Q 324  and Q 326  is supplied to one of the balance input terminals of the phase-shift circuit  36 Q via the transistor Q 392 . Also, the intermediate frequency signal −SIFQ extracted from the transistors Q 325  and Q 327  is supplied to the other one of the balance input terminals of the phase-shift circuit  36 Q via the transistor Q 393 .  
         [0075]     However, based on the control signal SSW, when the transistor Q 396  is on and the transistor Q 395  is off, the transistors Q 391  and Q 394  are turned on, and the transistors Q 392  and Q 393  are turned off. Therefore, the intermediate frequency signal +SIFQ extracted from the transistors Q 324  and Q 326  is supplied to the other one of the balance input terminals of the phase-shift circuit  36 Q via the transistor Q 391 . Also, the intermediate frequency signal −SIFQ extracted from the transistors Q 325  and Q 327  is supplied to one of the balance input terminals of the phase-shift circuit  36 Q via the transistor Q 394 .  
         [0076]     Therefore, since the phase of the intermediate frequency signal SIFQ supplied to the phase-shift circuit  36 Q is inverted or non-inverted in accordance with the control signal SSW, the intermediate frequency signal SIF of the ensemble A or the ensemble B is output from the addition circuit  38 . In this case, since the phase of the intermediate frequency signal SIFQ need only be inverted or non-inverted by the switching circuit  39 , it is possible to quickly switch the ensemble.  
         [0077]     Although the phase of the intermediate frequency signal SIFI is kept fixed, the intermediate frequency signal SIFI output from the mixer circuit  32 I may be supplied to a phase-shift circuit  36 I via a switching circuit having the same construction as that of the switching circuit  39 , and the switching. circuit may be kept fixed.  
         [0078]      FIG. 5  shows a circuit  34 Q of a portion which switches the phase of the local oscillation signal SLQ within the phase processing circuit  34  in  FIG. 3 . That is, the mixer circuit  32 Q is formed as a double balance-type as described in  FIG. 4 , and the received signals A 1  to A 3  and B 1  to B 3  are extracted as a balanced type and are supplied to the transistors Q 322  and Q 323 .  
         [0079]     Furthermore, the switching circuit  34 Q is formed as a double balanced-type by the transistors Q 341  to Q 347 . The local oscillation signal +SLQ of one of the phases is supplied to the transistors Q 345  and Q 346 , and the local oscillation signal −SLQ of the other phases is supplied to the transistors Q 344  and Q 347 . Also, the balanced-type control signal SSW is supplied to the transistors Q 342  and Q 343 .  
         [0080]     Then, based on the control signal SSW, when the transistor Q 342  is on and the transistor Q 343  is off, the transistors Q 344  and Q 345  are turned on, and the transistors Q 346  and Q 347  are turned off. Therefore, the local oscillation signal +SLQ is supplied to the transistors Q 324  to Q 327  via the transistor Q 345  and further via the emitter-follower transistor Q 349 . Also, the local oscillation signal −SLQ is supplied to the transistors Q 325  and Q 326  via the transistor Q 344  and further via the emitter-follower transistor Q 348 .  
         [0081]     However, based on the control signal SSW, when the transistor Q 343  is on and the transistor Q 342  is off, the transistors Q 346  and Q 347  are turned on, and the transistors Q 344  and Q 345  are turned off. Therefore, the local oscillation signal +SLQ is supplied to the transistors Q 325  and Q 326  via the transistor Q 346  and further via the transistor Q 348 . Also, the local oscillation signal −SLQ is supplied to the transistors Q 324  and Q 327  via the transistor Q 347  and further via the transistor Q 349 .  
         [0082]     Therefore, since the phase of the local oscillation signal SLQ supplied to the mixer circuit  32 Q is made to lead or reversed in accordance with the control signal SSW, the intermediate frequency signal SIF of the ensemble A or the ensemble B is output from the addition circuit  38 . Also in this case, since the phase of the local oscillation signal SLQ need only be inverted or non-inverted by the switching circuit  34 Q, the ensemble can be switched quickly.  
         [0083]     Many different embodiments of the present invention may be constructed without departing from the spirit and scope of the present invention. It should be understood that the present invention is not limited to the specific embodiments described in this specification. To the contrary, the present invention is intended to cover various modifications and equivalent arrangements included within the spirit and scope of the invention as hereafter claimed. The scope of the following claims is to be accorded the broadest interpretation so as to encompass all such modifications, equivalent structures and functions.

Technology Category: 5