Patent Document

FIELD OF THE INVENTION 
     The present invention is generally related to Phase Lock Loop (PLL) circuits, and more particularly to PLL semiconductor integrated circuits having integrated voltage controlled oscillators (VCOs) such as those used in wireless communication systems. 
     BACKGROUND OF THE INVENTION 
     Phase Lock Loop (PLL) integrated circuits (ICs) find practical advantages in many electronic circuits, and in particular, in wireless communications systems dealing with high-speed data transfer including receivers. In wireless systems, it is critical to achieve both fast lock and perfect tuning of a voltage controlled oscillator (VCO) comprising a portion of the PLL. In conventional PLL circuits having an integrated VCO, tuning of the VCO may take a relatively long period of time which may not be tolerable in circuit designs handling high data transfer rates. The longer an oscillator takes to tune directly impacts the lock time of the PLL. Moreover, prior art semiconductor PLL circuits require a relatively large overhead circuit and extra power dissipation which are undesirable characteristics when embodied in silicon designs, such as in wideband code division multiple access (WCDMA) chipsets. 
     There is desired an improved PLL having both a fast lock time and an accurate self-tuning VCO having both a reduced overhead circuit and generating less power dissipation than those presently available. 
     SUMMARY OF THE INVENTION 
     The present invention achieves technical advantages as a fast lock/self-tuning VCO based PLL whereby counters used in a divider are monitored to determine the lock condition of the PLL. A digital to analog converter (DAC) controls the course tuning of the VCO and is predistorted to lineraize the tuning of the VCO. Advantageously, the present invention merges the DAC with the VCO. The present invention allows an almost perfect tuning of the VCO and fast lock operation, which is critical in wireless systems handling high speed data transfer, such as WDCMA based communications. The present invention is preferably implemented in RFSiGe or CMOS process in a WDCMA chipset, and can be used in other systems such as GSM and EDGE. The present invention is also advantageous for use in new fractional-N PLL products. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 depicts a PLL loop circuit according to the present invention whereby a controller monitors the state of counters forming a divider circuit; 
     FIG. 2 depicts a schematic of one implementation to merge a DAC with a VCO; 
     FIG. 3 depicts a graph of the oscillator frequency of the VCO as a function of the reverse voltage, depicting the non-linear relationship; 
     FIG. 4 illustrates graphs depicting how to make the oscillator frequency of the VCO function linear by predistorting the DAC to be a non-linear voltage generator having an operating curve being opposite to the effect of the varactor shown in FIG. 3; 
     FIG. 5 depicts a schematic illustrating another way to make to achieve linear control of the frequency by merging the DAC with the varactor; 
     FIG. 6 is a schematic of a weighted array using thermometer decoding for each code increment by mixing only one varactor such that the sizing of the varactor is made to have a linear relation; and 
     FIG. 7 depicts a binary search technique making the residue in the counters equal to zero, whereby the DAC is programmed by the binary search technique until the counter residue is zero or negligible. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     An interpolation technique of the first preferred embodiment of the present invention can be applied to both integer and fractional frequency synthesizers and is seen in FIG.  1 . 
     Initially, a PLL loop circuit 10 shown in FIG. 1 is opened and a fine-tuning input  12  to a VCO  14  is set to some fixed reference voltage (i.e. ½ supply voltage: VCC/2) and a coarse tuning input  16  is set to a minimum frequency i.e. a digital input  18  to a digital-to-analog converter (DAC)  20  is set to a digital code 00000 by a loop controller  22 . The counters A and B in divider 24 are controlled by controller  22  to then start to countdown and set the division ratio, where: 
       N=B ( P )+ A.   
     According to the present invention, the tuning controller  22  monitors the state of each counter A and B and its residue for some predetermined time window: 
     T st =M×F ref  where M is the number of cycles of monitoring. 
     Since F in  is smaller than the target output frequency, where the DAC  20  digital input is set to , for every F ref  cycle the counters A and B will not overflow. 
     For each T st  time iteration the counters will have a residue number called N r ,: 
     By neglecting the residue in counter A after M cycles, it can be proved that the error Δf 0  in frequency is: 
     
       
         Δ f   0 ( N   R )×F ref  thus, 
       
     
     
       
         Δ f   0   =ΔB×P×F   ref   
       
     
     Examples of this interpolation technique illustrate the advantages of the present invention. 
     EXAMPLE 1 
     
       
         
           
             Assuming 
              
             
                 
             
              
             a 
              
             
                 
             
              
             linear 
              
             
                 
             
              
             
               relation 
               : 
               
                 
                   
                     
                       DAC 
                        
                       
                           
                       
                        
                       
                         input 
                         : 
                         00000 
                       
                     
                   
                   
                     
                       
                         
                           F 
                           out 
                         
                          
                         
                             
                         
                          
                         
                           ( 
                           VCO 
                           ) 
                         
                       
                       : 
                       
                         855 
                          
                         
                             
                         
                          
                         MHz 
                       
                     
                   
                 
                 
                   
                     ⋮ 
                   
                   
                     ⋮ 
                   
                 
                 
                   
                     
                       DAC 
                        
                       
                           
                       
                        
                       
                         input 
                         : 
                         11111 
                       
                     
                   
                   
                     
                       
                         
                           F 
                           out 
                         
                          
                         
                             
                         
                          
                         
                           ( 
                           VCO 
                           ) 
                         
                       
                       : 
                       
                         1010 
                          
                         
                             
                         
                          
                         MHz 
                       
                     
                   
                 
               
             
           
         
                 
         
             
         
      
     
     With a 5-bit DAC: 
     K DAC =5 MHz/LSB where K DAC  is the VCO/DAC gain 
     Assuming a linear relation: 
     Referenced clock: 10 MHz 
     P=8 (prescalar number) 
     If want target frequency=880 MHz                F   ref     =     10                 MHz            
            F   0     =     880                 MHz            
          N   =         880                 MHz       10                 MHz       =     88   =         (     11   ×   8     )     +   0     =             〉          (       B   =   11       A   =   0       )                            
     Initially, the digital input at the DAC  20  input is set at , and therefore, f 0 =855 MHz. Now, to determine the correction of the input to the DAC  20  to get near the target frequency of 880 MHz, the Frequency error to be corrected is determined: 
     
       
         Δ f   0 =880 MHz−855 MHz=25 MHz 
       
     
     since Δf 0 =ΔB×P×F ref           Δ                 B     =         Δ                 fo       P   ×     F   ref         =         25                 MHz       8   ×   10                 MHz       =   0.3125                              
     which is difficult to detect. 
     If M=16 clock cycles, the monitoring time of the counter residue is 16 cycles of the reference clock. 
     
       
         Then: Δ B   m   =M×ΔB =16×0.3125=5 
       
     
     The residue number in the B counter is 5, and that is exactly what is needed to correct the DAC input to get the correct desired frequency: 
     In summary, the DAC input=0+5=5 [00101] to get 
     
       
         F out =855+5×(5 MHz/ LSB )=880 MHz  
       
     
     EXAMPLE 2 
     Desired target frequency: 960 MHz 
     Where: B=12 P=8 A=F ref =10 MHz 
     thus, frequency error Δf 0 =105 MHz          Δ                 B     =         105                 MHz       10                 MHz   ×   8       =   1.3125                            
     therefore: ΔB m =16×1.3125=21 
     Thus, the DAC digital input is corrected (increased) by 21 [10101] Hence: 
     
       
           f   0 =21×5 MHz/LSB+855=960 MHz 
       
     
     An example of not neglecting the residue in A in the equation N=P(B)+A, is illustrated in the next example: 
     EXAMPLE 3 
     
       
         Δ f   0   =ΔN×F   ref   
       
     
     for: F 0 =920 MHz F ref =10 MHz              N   =         (     11   ×   8     )     +     4                 where                 B                  =   11                 A              =   4                   Δ                 N     =         Δ                 fo       F   ref       =         920   -   855     10     =   6.5                 For                 16                 clock                 cycles                   (     M   =   16     )                   Δ                   N   M       =       16   ×   6.5     =   104               Δ                 B     =         Δ                   N   M       P     =       104   8     =   13                              
     Thus, the digital input to DAC  20  has to be incremented by 13 LSBs: 
     
       
           F   out =855+(13×5)=920 MHz 
       
     
     After the coarse tuning through the DAC  20 , a phase detector  26  which includes a charge pump, is enabled by controller  22 , the precharge circuit to Vref is disabled, and the PLL finishes the lock through fine tuning. Most of the lock time will be phase locking, after finishing the tuning. 
     One circuit implementation achieving the VCO tuning is shown in FIG. 2 at  30 , with the varactors depicted as diodes, but alternatively could also comprise MOS—type varactors. The oscillator frequency of the VCO function of the reverse voltage is not linear, and is shown in FIG.  3 . 
     Advantageously, according to the present invention, one way to make the oscillator frequency of the VCO function linear and get more accurate tuning is to make the DAC  20  a non-linear voltage generator with an opposite curve to that shown in FIG. 3, with the effect of the varactor operating as shown in FIG.  4 . 
     According to an alternative embodiment of the present invention, another way to make this linear control of the frequency through the DAC  20  is to merge the DAC with the varactor, as shown at  40  in FIG. 5 
     Capacitor C 0  is a function of (Vmin, Vmax), where          [           Vmin   →       C   0        min                 Vmax   →       C   0        max             ]                               
     For code=[code n  . . . code 0 ]=[00 . . . 0] where the capacitance is maximum and f 0  is minimum, by incrementing the code by 1 from minimum, C 0  is decreased, then capacitor C 1  is decreased, . . . and finally capacitor C n  is decreased. If capacitors C 0 , C 1  and . . . C n  are sized accordingly, then the relation: 
     f 0 =function (code) can be made almost linear, where sizing depends on the relation C (V). 
     However, since the array is binary weighted, then the linearity is difficult to meet. 
     So, another improvement according to the present invention is based on an un-weighted array using thermometer decoding, as shown at  50  in FIG.  6 . For each code increment, incrementing is achieved by mixing only one varactor. Advantageously, the sizing of the varactor can be made so to have a linear relation between f 0  and code. 
     This technique can even be applied to other types of VCOs. The present invention derives technical advantages by using the content of the counters to easily know if the VCO  14  is tuned. 
     For more accurate tuning, multiple adjustments of the tuning DAC  20  can be used instead of one adjustment, to overcome the problems associated with process variations and any non-linearity effects. 
     An example to illustrate this, similar to Example 3, is shown below. 
     EXAMPLE 4 
     Target frequency F 0 =920 MHz 
     F ref =10 MHz 
     N=11 (P)+4 (P:8 prescalar modulus 8/9) 
     Assume that the DAC gain is slightly different than 5 MHz/LSB, and in this example 5.5 MHz/LSB (process effect). [The 5.5 MHz/LSB is a prior not known] 
     From Example 1, ΔB=13 
     Using a two-step adjustment for the DAC, so: 
     The DAC  20  has to be incremented by 6 LSBS        (     ≤       1   2                   Δ                 B       )                          
     Then, using another times cycle of 13 reference cycles to measure the new ΔB: 
     
       
         Δ B   i =6 ΔA =3 
       
     
     Reference cycle:          Δ                 N     =         920   -   888     10     =   3.2                           Δ N   m =51.2≅51 
     Note, normally Δf=6 LSB=5.5=33 MHz: 
     In the second step, the DAC digital input is moved again by 6 LSBs. The final frequency is 921 MHz, which is around 0.1% of the target frequency. By increasing the dynamic range of the tuning DAC  20  and using more reference cycles for time, and also using ΔA information for DAC adjustment, more accuracy can be achieved. 
     However, the number of increments of DAC  20 , the number of reference cycles of calibration, and the number of adjustment steps (phases) depends on the allowed time of the calibration. An optimum design can be found in this case. For the previous example, 2 adjustment steps, 5-bit DAC, and 16 reference cycles for adjustment, needs at least 3.2 μs without accounting overhead times due to some timing set-up. 
     Advantageously, the overflow of the counters can be used to detect a large ΔB during tuning because ΔB max is limited by the programmed ΔB for a targeted change. 
     Advantages of this approach: 
     1. The VCO is tuned automatically when the PLL is powered on (or on any reset) and during the lock time. 
     2. The control line is precharged to almost the final setting (including charging the filter) and hence, achieving a fast lock time when switched from tuning to locking. 
     3. The tuning algorithm is very simple, and does not need a large overhead circuit and time. 
     4. The DAC is merged with the VCO, saving circuit silicon space and easing the tuning. 
     5. Pre-distortion of the DAC eases the tuning to almost a perfect situation. 
     6. The technique can be applied to any PLL type circuit, etc. 
     Another preferred method of performing self-turning will now be discussed. This alternate approach is based on a binary search technique to make the residue in the counter=0. So, the DAC is programmed by binary search technique until the counter residue is 0 or negligible, as shown at  60  in FIG.  7 . 
     This approach  60  of self-tuning is similar to a successive approximation technique used in analog-to-digital converters (ADCs). This binary search technique can also be combined with the interpolation technique described with reference to FIGS.  1 — 4  as an automatic tuning technique. This binary search technique  60  works very well, however, it requires at least N times the reference time to finish tuning, where N is the number of bits of the VCO DAC. To reduce the tuning time, initially, M bits out of N bits (M&lt;N) are tuned using the binary search technique, then with the interpolation technique  10  being used to determine the rest of the bits (N−M). The partition of bits depends on the allowed PLL lock time, and the accuracy of the interpolation technique. Of course, other binary search techniques can be used. 
     Though the invention has been described with respect to a specific preferred embodiment, many variations and modifications will become apparent to those skilled in the art upon reading the present application. It is therefore the intention that the appended claims be interpreted as broadly as possible in view of the prior art to include all such variations and modifications.

Technology Category: 5