Patent Document

BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The present invention relates to a transmission method and a transmitter using the Ultra Wide-Band (UWB) system for transmission. 
   2. Description of Related Art 
   Particular attention has been paid to the UWB system as one of wireless transmission systems. The UWB system realizes transmission using a very wide transmission band of, for example, several gigahertzes and using very short pulses. 
   A recent trend is to put SS (Spread Spectrum) based wireless LAN (Local Area Network) systems to practical use. There are proposed SS based UWB transmission systems for PAN applications and the like. 
   The SS systems include the DS (Direct Spread) system. According to this system, the transmission side multiplies an information signal by a random code sequence called a PN (Pseudo Noise) code to spread a dedicated band for transmission. The reception side multiplies the received spread information signal by the PN code to reversely spread the information signal for reproduction. 
   The UWB transmission system includes two types: DS-UWB and impulse-UWB. The DS-UWB system maximizes spread code speeds of DS information signals. The impulse-UWB system configures an information signal using an impulse signal sequence having a very short cycle of approximately several hundred picoseconds to send and receive the signal sequence. 
   The DS-UWB can control spectra using PN code speeds, but needs to fast operate logic circuits in units of GHz. The power consumption increases dramatically. On the other hand, the impulse-UWB system can be configured in combination with a pulse generator and a low-speed logic circuit. There is an advantage of decreasing the current consumption. However, the pulse generator makes it difficult to control spectra. 
   Both systems implement high-speed data transmission by spreading signals to an ultra-high frequency band, e.g., between 3 and 10 GHz for transmission and reception. The dedicated bandwidth is expressed in units of GHz so that a value approximate to 1 results from division of the dedicated bandwidth by a center frequency (e.g., 1 to 10 GHz). The dedicated bandwidth is ultra wide compared to bandwidths normally used for wireless LANs based on the W-CDMA or cdma2000 system, and the SS (Spread Spectrum) or OFDM (Orthogonal Frequency Division Multiplexing) system. 
   Since the impulse-UWB system uses a very narrow pulse for the impulse signal, a very wide band is used in terms of the frequency spectrum. Consequently, an input information signal merely indicates a power smaller than the noise level in respective frequency domains. Available modulation systems include PPM (Pulse Position Modulation) to represent a code according to a position between pulses, Bi-phase Modulation to represent a code according to a pulse&#39;s phase change, and amplitude modulation. 
     FIG. 10  shows a configuration example of a conventional UWB transceiver. An antenna  11  is connected to an antenna changer  13  via a band-pass filter  12 . The antenna changer  13  is connected to reception-related circuits and transmission-related circuits. The antenna changer  13  functions as a selection switch to operate in interlock with transmission and reception timings. The band-pass filter  12  passes signals of transmission bandwidths of several gigahertzes such as 4 to 9 GHz used for the system. 
   The reception-related circuits connected to the antenna changer  13  include a low noise amplifier  14 , 2-system multipliers  15 I and  15 Q, low pass filters  16 I and  16 Q, and analog-digital converters  17 I and  17 Q. The low noise amplifier  14  amplifies an output from the antenna changer  13  for reception. The multipliers  15 I and  15 Q multiply an output from the low noise amplifier  14  by outputs from pulse generators  25 I and  25 Q. The low pass filters  16 I and  16 Q eliminate high frequency components from outputs from the multipliers  15 I and  15 Q. The analog-digital converters  17 I and  17 Q sample outputs from the low pass filters  16 I and  16 Q. 
   Output pulses from the pulse generator  25 I and  25 Q are phase-shifted from each other by the specified amount. The analog-digital converter  17 I samples I-channel transmission data. The analog-digital converter  17 Q samples Q-channel transmission data. Received data for each channel is supplied to the baseband circuit  30  for reception processing. In this example, received data for the I channel is used as is. Received data for the Q channel is used as an error signal. 
   As transmission-related circuits, the multiplier  26  is supplied with transmission data output from the baseband circuit  30 . The transmission data is multiplied by an output from the pulse generator  25 I. The transmission data output from the baseband circuit  30  is modulated, e.g., as an NRZ (Non Return to Zero) signal. The multiplier  26  multiplies the transmission data by an output from the pulse generator  25 I to generate a bi-phase modulated pulse. This becomes a signal modulated by the so-called BPSK (Binary Phase Shift Keying) system. In order to allow the pulse generator  25 I to generate pulses, there is provided a Voltage Controlled Temperature Compensated Crystal Oscillator (VCTCXO, hereafter simply referred to as an oscillator)  21  to control oscillation frequencies of the oscillator  21  based on an error signal acquired from received data for the Q channel, for example. 
   An oscillation signal from the oscillator  21  is supplied to a PLL (phase locked loop) circuit  22 . A voltage control oscillator  23  constitutes a loop for the PLL circuit  22 . An oscillated output from the voltage control oscillator  23  is supplied to the pulse generator  25 I to generate a pulse synchronized to the oscillated output from the oscillator  23 . A phase shifter  24  supplies a pulse generator  25 Q with an output from the oscillator  23  by delaying a specified cyclic phase. This makes it possible to generate a short wavelength pulse synchronized with the oscillated output from the oscillator  23  at a timing delayed from an output pulse of the pulse generator  25 I. 
   A multiplier  26  multiplies an output pulse from the pulse generator  25 Q by the transmission data to use the multiplication output as a transmission signal. The transmission signal output from the multiplier  26  is supplied to a power amplifier  27  and is amplified there for transmission. The amplified output is supplied to the band-pass filter  12  via the antenna changer  13 . The band-pass filter  12  limits the band to pass only signals for the transmission band. The transmission signal is then transmitted from the antenna  11 . 
   Non-patent document 1 outlines the UWB system. 
   [Non-patent document 1] 
   Nikkei Electronics, 11 Mar. 2002, pp. 55–66. 
   A pulse used for the impulse-UWB system is a signal having the wideband frequency spectrum. The time domain is equivalent to a monocycle waveform expressed by equation 1, for example. 
   
     
       
         
           
             
               
                 
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   In equation 1, tP represents the time from the monocycle waveform center to a peak value. In the case of tP=200 [psec], for example, the time waveform becomes a monocycle waveform generated at its maximum value of ±200 [psec] as shown in  FIG. 11 . We can confirm that the monocycle waveform&#39;s spectrum has the maximum value of approximately 1 [GHz] and the −3 dB bandwidth of approximately 1 [GHz]. 
   We examine generating a single sideband of the monocycle waveform for frequency conversion. The reason is that the UWB system specifies the following two spectrum requirements for transmission pulses. 
   (1) The US FCC spectrum mask specification, one of UWB specifications, requires that radiation levels be decreased in the bands except 3.1 through 10.6 [GHz]. 
   (2) The band of 4.9 through 5.8 [GHz] contains 5 GHz wireless LANs that should be avoided. 
   In consideration for these requirements, we can assume to be able to solve the above-mentioned problems of the spectrum in the UWB communication system as follows. That is to say, the spectrum in  FIG. 12  is converted into the frequency range, e.g., between 3.1 and 4.9 [GHz] to generate an upper side band spectrum as shown in  FIG. 13 . There is provided a method of frequency converting the monocycle waveform in the upper side band. The method subtracts a signal obtained as a product of multiplying a Hilbert transformed monocycle waveform in  FIG. 14  by a 3.1 [GHz] sine carrier from a signal obtained as a product of multiplying the monocycle waveform by a 3.1 [GHz] cosine carrier. 
   A pulse waveform in  FIG. 15  represents the time waveform resulting from the spectrum in  FIG. 12 . The envelope&#39;s amplitude gradually increases, peaks at the origin, and gradually decreases. Accordingly, it can be understood that the envelope approximates to a triangle. Further, it can be understood that a 6-cycle pulse waveform constitutes major amplitude components. 
   To solve the above-mentioned problems of the spectrum in the UWB communication system, we can come to a solution generate an N-cycle pulse whose envelope is amplified and is formed as a triangle. For example, the waveform in  FIG. 15  has the duration of approximately 2 [nsec]. Arranging this pulse waveform in a series enables the BPSK communication at 500 [Mb/s] by preventing a series of pulse waveforms from overlapping with each other. 
   To achieve a higher communication rate such as 1 [Gb/s], however, the waveform in  FIG. 15  needs to be arranged at a 1 [nsec] interval. Consequently, some waveforms may overlap with each other. When the receiver uses a band-pass filter, it is known that an impulse response of the band-pass filter causes a previous pulse&#39;s amplitude to affect the subsequent pulses. This problem is called an inter-symbol interference and should be considered when narrowing the band for improving the frequency utilization. 
   According to the Nyquist&#39;s theorem, a baseband bandwidth of ½T [Hz] is required to transmit pulses at a T [sec] interval without distortion, where 1/T [Hz] is the Nyquist bandwidth. Since the frequency under discussion ranges from 3.1 to 4.9 [GHz], the bandwidth is 1.8 [GHz] and the baseband bandwidth is its half, i.e., 900 [MHz]. It fully ensures the minimum baseband bandwidth of 500 [MHz] to transmit pulses at a T [sec] interval but is 10[%] fall short of the 1 [GHz] Nyquist bandwidth. 
   Nyquist showed that the Nyquist filter should be used to satisfy the condition of no distortion below the Nyquist bandwidth. However, it is difficult to create a 1 [GHz] baseband digital filter. The reason is that creating the intended digital filter requires, e.g., an 8-bit D/A converter operating at least at a sampling frequency of approximately 4 [GHz]. Presently, there is a marketed example as a standalone unit that uses four D/A converters at 1.25 [Gsamples/sec] to acquire 5 [Gsamples/sec] Though such product is available on the current technological level, the design is unfavorable from the viewpoint of the cost effectiveness between the power consumption and installation costs when the UWB communication is applied to the consumer equipment. 
   An alternative to the baseband digital filter may be a band-pass filter (BPF) for high frequency bands.  FIG. 16  shows an impulse response when a 5-polar Butterworth filter is used for the BPF having the 4 [GHz] center frequency and the 1.8 [GHz]band. As seen from the impulse response in  FIG. 16 , its main wave in the vicinity of 0.8 [nsec] indicates that the BPF is subject to a delay time of 0.8 [nsec]. However, there is also generated a swell as large as one third of the main wave in the vicinity of 1.8 [nsec]. In this manner, an inter-symbol interference occurs due to the impulse response when a non-Nyquist filter is used. 
   SUMMARY OF THE INVENTION 
   The present invention has been made in consideration of the foregoing. It is therefore an object of the present invention to provide a high communication speed by performing pulse modulation and generating transmission signals so as to allow for an effect of the inter-symbol interference resulting from the use of non-Nyquist filters in the UWB communication. 
   The present invention generates a reference clock signal; sequentially outputs spread data at a specified timing synchronized with the reference clock, wherein the spread data results from directly spreading transmission data with a spreading code; distributes the spread data into two sequences of data at a specified timing synchronized with the reference clock; generates first and second pulse shaping signals at a specified timing synchronized with the reference clock; generates a cosine carrier and a sine carrier; multiplies one of the two sequences of data, the first pulse shaping signal, and the cosine carrier together; multiplies the other of the two sequences of data, the second pulse shaping signal, and the sine carrier together; and synthesizes outputs from the multiplications to acquire an output signal for transmission. 
   The present invention distributes spread data directly distributed by a spreading code sequence into two sequences of transmission data in synchronization with the reference clock signal. When the two sequences of transmission data are assumed to be I and Q data maintaining the time relationship so that their transition timings shift by a half cycle, this signal becomes an NRZ signal having a signal rate that is half the spread data. The cosine carrier and the sine carrier are orthogonally phase-shifted by 90 degrees. 
   One of two sequences of transmission data is multiplied by the first pulse shaping signal and the cosine carrier to generate a BPSK modulated I pulse. The other of two sequences of transmission data is multiplied by the second pulse shaping signal and the sine carrier to generate a BPSK modulated Q pulse. The BPSK modulated I and Q pulses are synthesized to be an output signal. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a configuration diagram showing the pulse phase modulation configuration of a transmitter according to a first embodiment of the present invention; 
       FIG. 2  is a waveform chart exemplifying signal waveforms according to the configuration in  FIG. 1 ; 
       FIG. 3  is a configuration diagram showing the pulse phase modulation configuration of a transmitter according to a second embodiment of the present invention; 
       FIG. 4  is a waveform chart exemplifying signal waveforms according to the configuration in  FIG. 3 ; 
       FIG. 5  is an explanatory diagram showing the constellation of baseband signals according to the configuration in  FIG. 3 ; 
       FIG. 6  is a configuration diagram showing the pulse phase modulation configuration of a transmitter according to a third embodiment of the present invention; 
       FIG. 7  is a waveform chart exemplifying signal waveforms according to the configuration in  FIG. 6 ; 
       FIG. 8  is a block diagram exemplifying the configuration of a pulse shaping signal generation circuit; 
       FIG. 9  is a waveform chart exemplifying signal waveforms of the circuit in  FIG. 8 ; 
       FIG. 10  is a block diagram exemplifying the configuration of a communication apparatus according to the conventional UWB system; 
       FIG. 11  is a waveform chart showing a mono-cycle pulse; 
       FIG. 12  is a frequency characteristics diagram showing a frequency spectrum of the mono-cycle pulse in  FIG. 11 ; 
       FIG. 13  is a frequency characteristics diagram showing a frequency spectrum of the upper wave generated in the mono-cycle pulse in  FIG. 11 ; 
       FIG. 14  is a waveform chart showing a waveform generated by Hilbert transforming the mono-cycle pulse in  FIG. 11 ; 
       FIG. 15  is a waveform chart showing a time waveform for the upper side band of the mono-cycle pulse in  FIG. 13 ; and 
       FIG. 16  is a characteristics diagram showing an impulse response of the band-pass filter. 
   

   DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
   The first embodiment of the present invention will be described in further detail with reference to  FIGS. 1 and 2 . 
   The embodiment provides the configuration for processing transmission signals of a transmitter or a transceiver applied to the UWB system for wireless transmission.  FIG. 1  shows the pulse phase modulation configuration of a transmitter  100  according to the embodiment. The transmitter  100  is provided with a first oscillator  101  that outputs a reference clock S 101 . 
   Further, there are provided a spread data output section  103  and a demultiplexer  104 . The spread data output section  103  directly spreads transmission data STXD using a specified spreading code SSS. The demultiplexer  104  distributes spread data S 103  generated by the direct spreading into I data S 104 A and Q data S 104 B. Moreover, there are provided an I-pulse shaping signal generator  105  and a Q-pulse shaping signal generator  106 . The I-pulse shaping signal generator  105  generates an I-pulse shaping signal S 105  based on the reference clock S 101  from the first oscillator  101 . The Q-pulse shaping signal generator  106  generates a Q-pulse shaping signal S 106 . Specific configuration examples of the I-pulse shaping signal generator  105  and the Q-pulse shaping signal generator  106  will be described with reference to  FIGS. 8 and 9 . 
   Furthermore, there are provided an oscillator  102  and a phase shifter  107 . The oscillator  102  generates a carrier signal S 102 . The phase shifter  107  shifts a phase of the carrier signal S 102  and outputs a cosine carrier S 107 A and a sine carrier S 107 B. 
   The first multiplier  108  multiplies the cosine carrier S 107 A output from the phase shifter  107  by the I-pulse shaping signal S 105  output from the I-pulse shaping signal generator  105  to generate an I pulse S 108 . 
   The second multiplier  109  multiplies the sine carrier S 107 B output from the phase shifter  107  by the Q-pulse shaping signal S 106  output from the Q-pulse shaping signal generator  106  to generate an Q pulse S 109 . 
   An I pulse S 108  output from the first multiplier  108  is supplied to a third multiplier  110 . The multiplier  110  multiplies the I pulse S 108  by I data S 104 A output from the demultiplexer  104  to BPSK modulate the I pulse S 108  and generates an I pulse S 110 . 
   Likewise, a Q pulse S 109  output from the second multiplier  109  is supplied to a fourth multiplier  111 . The multiplier  111  multiplies the Q pulse S 109  by Q data S 104 B output from the demultiplexer  104  to BPSK modulate the Q pulse S 109  and generates a Q pulse S 111 . 
   There is provided an adder  112  that synthesizes the BPSK modulated I pulse S 110  with the BPSK modulated Q pulse S 111  to generate an output signal S 112 . To synthesize the pulses and generate the output signal S 112 , the adder  112  uses polarities for subtracting the BPSK modulated Q pulse S 111  from the BPSK modulated I pulse S 110 . For example, the output signal S 112  can be amplified for transmission and wirelessly transmitted as a UWB signal from an antenna. 
     FIG. 2  exemplifies signal waveforms that are processed by the components of the configuration in  FIG. 1 . The following describes the waveforms in  FIG. 2 . 
   The reference clock S 101  of  FIG. 2(   a ) is a 4 [GHz] clock. 
   Data CLK of  FIG. 2(   b ) is a 1 [GHz] clock signal. A configuration to supply the data clock is omitted from  FIG. 1 . 
   The spread data S 103  of  FIG. 2(   c ) is a 1 [Gbps] digital signal generated by directly spreading the transmission data STXD with the spreading code SSS. 
   The I data S 104 A of  FIG. 2(   d ) is a 500 [Mbps] NRZ signal generated by distributing the spread data S 103  in the demultiplexer  104  in the order indicated by broken lines. 
   The Q data S 104 B of  FIG. 2(   e ) is a 500 [Mbps] NRZ signal generated by distributing the spread data S 103  in the demultiplexer  104  in the order indicated by dot-dash lines. 
   The I-pulse shaping signal S 105  of  FIG. 2(   f ) is a stepped triangle-wave analog signal generated in the I-pulse shaping signal generator  105  and synchronizes with the I data S 104 A. 
   The Q-pulse shaping signal S 106  of  FIG. 2(   g ) is a stepped triangle-wave analog signal generated in the Q-pulse shaping signal generator  106  and synchronizes with the Q data S 104 B. 
   The cosine carrier S 107 A of  FIG. 2(   h ) is a 4 [GHz] NRZ signal. 
   The sine carrier S 107 B of  FIG. 2(   i ) is a 4 [GHz] NRZ signal and is phase-shifted from the cosine carrier S 107 A by 90 degrees. 
   The I pulse S 108  of  FIG. 2(   j ) results from multiplying the I-pulse shaping signal S 105  and the cosine carrier S 107 A together. 
   The Q pulse S 109  of  FIG. 2(   k ) results from multiplying the Q-pulse shaping signal S 106  and the cosine carrier S 107 B together. 
   The BPSK modulated I pulse S 110  of  FIG. 2(   l ) results from multiplying the I data S 104 A and the I pulse S 108  together. 
   The BPSK modulated Q pulse S 111  of  FIG. 2(   m ) results from multiplying the Q data S 104 B and the Q pulse S 109  together. 
   The output signal S 112  of  FIG. 2(   n ) is generated by subtracting the BPSK modulated Q pulse S 111  from the BPSK modulated I pulse S 110 . 
   The spread data output section  103  sequentially outputs the spread data S 103  in the pulse phase conversion configuration of the transmitter  100  according to the embodiment. The spread data S 103  synchronizes with the reference clock signal S 101  and is generated by directly spreading the transmission data STXD with the spreading code sequence SSS. The demultiplexer  104  divides the spread data S 103  into the I data S 104 A and the Q data S 104 B. 
   In this example, as seen from  FIGS. 2(   d ) and  2 ( e ), the I data S 104 A and the Q data S 104 B maintain the time relationship so that their transition timings shift by a half cycle. Both data are NRZ signals having a signal rate that is half the spread data S 103 . The I-pulse shaping signal generator  105  and the Q-pulse shaping signal generator  106  generate the I-pulse shaping signal S 105  ( FIG. 2(   f )) and the Q-pulse shaping signal S 106  ( FIG. 2(   g )) as stepped triangle waves. The I-pulse shaping signal S 105  and the Q-pulse shaping signal S 106  are generated at the timings so that each peak value of the waveforms corresponds to the center of the I data S 104 A and the Q data S 104 B, respectively. The configuration to generate a stepped triangle wave will be described later. 
   The first multiplier  108  multiplies cosine carrier S 107 A ( FIG. 2(   h )) by the I-pulse shaping signal S 105  ( FIG. 2(   f )) to generate the 6-cycle I pulse S 108  ( FIG. 2(   j )). The second multiplier  109  multiplies the sine carrier S 107 B ( FIG. 2(   i )) by the Q-pulse shaping signal S 106  ( FIG. 2(   g )) to generate the 6-cycle Q pulse S 109  ( FIG. 2(   k )). The cosine carrier S 107 A and the sine carrier S 107 B are orthogonal to each other with a phase difference of 90 degrees. Further, the third multiplier  110  BPSK modulates the I pulse S 108  ( FIG. 2(   j )) with the I data S 104 A ( FIG. 2(   d )). The fourth multiplier  111  BPSK modulates the Q pulse S 109  ( FIG. 2(   k )) with the Q data S 104 B ( FIG. 2(   e )). The adder  112  uses polarities for subtracting the BPSK modulated Q pulse S 111  ( FIG. 2(   m )) from the BPSK modulated I pulse S 110  ( FIG. 2(   l )) to synthesize both pulses and generates the output signal S 112  ( FIG. 2(   n )). 
   The output signal processed as mentioned above shows a constant envelope because the BPSK modulated I and Q pulses generate the timings whose time relationship is characterized by a half-cycle shift. Each of the I and Q pulses is an N-cycle pulse in itself. Accordingly, these pulses are considered to mediate between the DS-UWB system and the impulse-UWB system in terms of the UWB transmission systems. The constant envelope in the output signal makes it possible to use nonlinear amplifiers such as C-class amplifiers, realizing a transmitter with high power efficiency. 
   As mentioned above, the N-cycle pulse has the narrow-band frequency spectrum. Therefore, the following advantages are provided. No band-pass filters are needed for transmission. The transmission side is free from an inter-symbol interference. The I and Q pules are subject to little, if any, interference therebetween due to a career&#39;s orthogonal phase error. The reason is that both pulses maintain the time relationship with their timings shifted by a half cycle. No signal occurs at the timings. The signal rate of the I and Q pulses each is half the spread data. This doubles an interval of the I and Q pulses in themselves. The pulses are subject to little inter-symbol interference occurring in the band-pass filter at the reception side. 
   Consequently, the transmitter configuration according to the embodiment can decrease effects of inter-symbol interference due to the use of non-Nyquist filters and increase the UWB communication speed. 
   Referring now to  FIGS. 3 through 5 , the second embodiment of the present invention will be described. The same parts or components are depicted by the same reference numerals with reference to  FIGS. 3 through 5  for the second embodiment and  FIGS. 1 and 2  for the above-mentioned first embodiment. 
   Like the first embodiment, the second embodiment provides the configuration for processing transmission signals of a transmitter or a transceiver applied to the UWB system for wireless transmission.  FIG. 3  shows the pulse phase modulation configuration of a transmitter  120  according to the embodiment. The transmitter  120  comprises the first oscillator  101 , the oscillator  102 , the spread data output section  103 , the demultiplexer  104 , the I-pulse shaping signal generator  105 , the Q-pulse shaping signal generator  106 , and the phase shifter  107 . The fist oscillator  101  outputs the reference clock S 101 . The oscillator  102  generates the carrier signal S 102 . The spread data output section  103  directly spreads transmission data STXD using a specified spreading code SSS. The demultiplexer  104  distributes the spread data S 103  generated by the direct spreading into the I data S 104 A and the Q data S 104 B. The I-pulse shaping signal generator  105  generates the I-pulse shaping signal S 105 . The Q-pulse shaping signal generator  106  generates the Q-pulse shaping signal S 106 . The phase shifter  107  shifts a phase of the carrier signal S 102  and outputs the cosine carrier S 107 A and the sine carrier S 107 B. The configurations of these processing means are the same as those described in the first embodiment with reference to  FIG. 1 . Specific configuration examples of the I-pulse shaping signal generator  105  and the Q-pulse shaping signal generator  106  will be described later. 
   In this example, there is provided a fifth multiplier  121 . It multiplies the I data S 104 A output from the demultiplexer  104  by the I-pulse shaping signal S 105  output from the I-pulse shaping signal generator  105 . The multiplier  121  outputs a baseband signal S 121 . There is also provided a sixth multiplier  122 . It multiplies the Q data S 104 B output from the demultiplexer  104  by the Q-pulse shaping signal S 106  output from the shaping signal generator  106 . The multiplier  122  output s baseband signal S 122 . 
   Further, there are provided a third multiplier  110 , a fourth multiplier  111 , and the adder  112 . The third multiplier  110  multiplies an I baseband signal S 121  by the cosine carrier S 107 A to output the BPSK modulated I pulse S 110 . The fourth multiplier  111  multiplies a Q baseband signal S 122  by the sine carrier S 107 B to output the BPSK modulated Q pulse S 111 . The adder  112  synthesizes the BPSK modulated I pulse S 110  with the BPSK modulated Q pulse S 111  to generate the output signal S 112 . To synthesize the pulses and generate the output signal S 112 , the adder  112  uses polarities for subtracting the BPSK modulated Q pulse S 111  from the BPSK modulated I pulse S 110 . 
     FIG. 4  exemplifies signal waveforms that are processed by the components of the configuration in  FIG. 3 . The following describes the waveforms in  FIG. 4 . 
   The reference clock S 101  of  FIG. 4(   a ) is a 4 [GHz]clock. 
   Data CLK of  FIG. 4(   b ) is a 1 [GHz] clock signal. A configuration to supply the data clock is omitted from  FIG. 3 . 
   The spread data S 103  of  FIG. 4(   c ) is a 1 [Gbps] digital signal generated in the spread data output section  103  by directly spreading the transmission data STXD with the spreading code SSS. 
   The I data S 104 A of  FIG. 4(   d ) is a 500 [Mbps] NRZ signal generated by distributing the spread data S 103  in the demultiplexer  104  in the order indicated by broken lines. 
   The Q data S 104 B of  FIG. 4(   e ) is a 500 [Mbps] NRZ signal generated by distributing the spread data S 103  in the demultiplexer  104  in the order indicated by dot-dash lines. 
   The I-pulse shaping signal S 105  of  FIG. 4(   f ) is a stepped triangle-wave analog signal generated in the I-pulse shaping signal generator  105  and synchronizes with the I data S 104 A. 
   The Q-pulse shaping signal S 106  of  FIG. 4(   g ) is a stepped triangle-wave analog signal generated in the Q-pulse shaping signal generator  106  and synchronizes with the Q data S 104 B. 
   The I baseband signal S 121  of  FIG. 4(   h ) results from multiplying the I data S 104 A by the I-pulse shaping signal S 105 . 
   The Q baseband signal S 122  of  FIG. 4(   i ) results from multiplying the Q data S 104 B by the Q-pulse shaping signal S 106 . 
   The cosine carrier S 107 A of  FIG. 4(   j ) is a 4 [GHz] NRZ signal. 
   The sine carrier S 107 B of  FIG. 4(   k ) is a 4 [GHz] NRZ signal and is phase-shifted from the cosine carrier S 107 A by 90 degrees. 
   The BPSK modulated I pulse S 110  of  FIG. 4(   l ) results from multiplying the I baseband signal S 121  and the cosine carrier S 107 A together. 
   The BPSK modulated I pulse S 110  of  FIG. 4(   m ) results from multiplying the Q baseband signal S 122  and the sine carrier S 107 B together. 
   The output signal S 112  of  FIG. 4(   n ) is generated by subtracting the BPSK modulated Q pulse S 111  from the BPSK modulated I pulse S 110  and becomes the same as that in the first embodiment. 
   The spread data output section  103  sequentially outputs the spread data S 103  in the transmitter  120  according to the second embodiment. The spread data S 103  synchronizes with the reference clock signal S 101  and is generated by directly spreading the transmission data STXD with the spreading code sequence SSS. The demultiplexer  104  divides the spread data S 103  into the I data S 104 A and the Q data S 104 B. Here, the I data S 104 A and the Q data S 104 B maintain the time relationship so that their transition timings shift by a half cycle. Both data are NRZ signals having a signal rate that is half the spread data S 103 . 
   The I-pulse shaping signal generator  105  and the Q-pulse shaping signal generator  106  generate the I-pulse shaping signal S 105  and the Q-pulse shaping signal S 106  as stepped triangle waves. The I-pulse shaping signal S 105  and the Q-pulse shaping signal S 106  are generated at the timings so that each peak value of the waveforms corresponds to the center of the I data S 104 A and the Q data S 104 B, respectively. The fifth multiplier  121  multiplies the I data S 104 A and the I-pulse shaping signal S 105  together to generate the I baseband signal S 121 . The sixth multiplier  122  multiplies the Q data S 104 B and the Q-pulse shaping signal S 106  together to generate the Q baseband signal S 122 . 
   The third multiplier  110  multiplies the cosine carrier S 107 A and the I baseband signal S 121  together to output the BPSK modulated I pulse S 110 . The fourth multiplier  111  multiplies the sine carrier S 107 B and the Q baseband signal S 122  together to output the BPSK modulated Q pulse S 111 . The adder  112  uses polarities for subtracting the BPSK modulated Q pulse S 111  from the BPSK modulated I pulse S 110  to synthesize both pulses and generates the output signal S 112 . 
   The output signal S 112  generated in this manner is essentially the same as that described in the first embodiment. That is to say, the output signal shows a constant envelope and can realize a transmitter with high power efficiency. No band-pass filters are needed for transmission. The transmission side is free from an inter-symbol interference. The signal rate of the I and Q pulses each is half the spread data. This doubles an interval of the I and Q pulses in themselves. The pulses are subject to little inter-symbol interference occurring in the band-pass filter at the reception side. 
     FIG. 5  shows a constellation display of the I baseband signal S 121  and the Q baseband signal S 122 . As can be seen from  FIG. 5 , the constellation follows the state transition indicated by arrows between four symbol points on the I and Q axes. Since the state transition rotates 90 degrees at each symbol point, the modulation can be categorized as the π/2-shift BPSK (Binary Phase Shift Keying). 
   Referring now to  FIGS. 6 and 7 , the third embodiment of the present invention will be described. The same parts or components are depicted by the same reference numerals with reference to  FIGS. 6 and 7  for the third embodiment and  FIGS. 1 through 5  for the above-mentioned first and second embodiments. 
     FIG. 6  shows the pulse phase modulation configuration of a transmitter  130  according to the third embodiment of the present invention. The transmitter  130  comprises the first oscillator  101 , the oscillator  102 , the spread data output section  103 , the demultiplexer  104 , the I-pulse shaping signal generator  105 , the Q-pulse shaping signal generator  106 , and the phase shifter  107 . The fist oscillator  101  outputs the reference clock S 101 . The oscillator  102  generates the carrier signal S 102 . The spread data output section  103  directly spreads transmission data STXD using a specified spreading code SSS. The demultiplexer  104  distributes the spread data S 103  generated by the direct spreading into the I data S 104 A and the Q data S 104 B. The I-pulse shaping signal generator  105  generates the I-pulse shaping signal S 105 . The Q-pulse shaping signal generator  106  generates the Q-pulse shaping signal S 106 . The phase shifter  107  shifts a phase of the carrier signal S 102  and outputs the cosine carrier S 107 A and the sine carrier S 107 B. The configurations of these processing means are the same as those described in the first and second embodiments with reference to  FIG. 3 . Specific configuration examples of the I-pulse shaping signal generator  105  and the Q-pulse shaping signal generator  106  will be described later. 
   In this example, there are provided a seventh multiplier  131  and an eighth multiplier  132 . The multiplier  131  multiplies the I data S 104 A output from the demultiplexer  140  and a cosine carrier output from the phase shifter  107  to output the BPSK modulated cosine carrier S 131 . The multiplier  132  multiplies the Q data S 104 B output from the demultiplexer  140  and a sine carrier output from the phase shifter  107  to output the BPSK modulated sine carrier S 132 . 
   Further, there are provided the third multiplier  110 , the fourth multiplier  111 , and the adder  112 . The multiplier  110  multiplies the cosine carrier S 131  BPSK-modulated by the multiplier  131  and the I-pulse shaping signal S 105  output from the I-pulse shaping signal generator  105  together to output the BPSK modulated I pulse S 110 . The multiplier  111  multiplies the sine carrier S 132  BPSK-modulated by the multiplier  132  and the Q-pulse shaping signal S 106  output from the Q-pulse shaping signal generator  106  together to output the BPSK modulated Q pulse S 111 . The adder  112  synthesizes the I pulse S 110  BPSK-modulated by the third multiplier  110  with the Q pulse S 111  BPSK-modulated by the fourth multiplier  111  to generate the output signal S 112 . To synthesize the pulses and generate the output signal S 112 , the adder  112  uses polarities for subtracting the BPSK modulated Q pulse S 111  from the BPSK modulated I pulse  FIG. 7  exemplifies signal waveforms that are processed by the components of the configuration in  FIG. 6 . The following describes the waveforms in  FIG. 7 . 
   The reference clock S 101  of  FIG. 7(   a ) is a 4 [GHz] clock. 
   Data CLK of  FIG. 7(   b ) is a 1 [GHz] clock signal. A configuration to supply the data clock is omitted from  FIG. 6 . 
   The spread data S 103  of  FIG. 7(   c ) is a 1 [Gbps] digital signal generated in the spread data output section  103  by directly spreading the transmission data STXD with the spreading code SSS. 
   The I data S 104 A of  FIG. 7(   d ) is a 500 [Mbps] NRZ signal generated by distributing the spread data S 103  in the demultiplexer  104  in the order indicated by broken lines. 
   The Q data S 104 B of  FIG. 7(   e ) is a 500 [Mbps] NRZ signal generated by distributing the spread data S 103  in the demultiplexer  104  in the order indicated by dot-dash lines. 
   The I-pulse shaping signal S 105  of  FIG. 7(   f ) is a stepped triangle-wave analog signal generated in the I-pulse shaping signal generator  105  and synchronizes with the I data S 104 A. 
   The Q-pulse shaping signal S 106  of  FIG. 7(   g ) is a stepped triangle-wave analog signal generated in the Q-pulse shaping signal generator  106  and synchronizes with the Q data S 104 B. 
   The cosine carrier S 107 A of  FIG. 7(   h ) is a 4 [GHz] NRZ signal. 
   The sine carrier S 107 B of  FIG. 7(   i ) is a 4 [GHz] NRZ signal and is phase-shifted from the cosine carrier S 107 A by 90 degrees. 
   The BPSK modulated cosine carrier S 131  of  FIG. 7(   j ) results from multiplying the cosine carrier S 107 A and the I data S 104 A together. 
   The BPSK modulated sine carrier S 132  of  FIG. 7(   k ) results from multiplying the cosine carrier S 107 B and the I data S 104 B together. 
   The BPSK modulated I pulse S 110  of  FIG. 7(   l ) results from multiplying the BPSK modulated cosine carrier S 131  and the I-pulse shaping signal S 105  together. 
   The BPSK modulated I pulse S 111  of  FIG. 7(   m ) results from multiplying the BPSK modulated sine carrier S 132  and the Q-pulse shaping signal S 106  together. 
   The output signal S 112  of  FIG. 7(   n ) is generated by subtracting the BPSK modulated Q pulse S 111  from the BPSK modulated I pulse S 110  and becomes the same as that in the first and second embodiments. 
   The spread data output section  103  sequentially outputs the spread data S 103  in the transmitter  120  according to the third embodiment. The spread data S 103  synchronizes with the reference clock signal S 101  and is generated by directly spreading the transmission data STXD with the spreading code sequence SSS. The demultiplexer  104  divides the spread data S 103  into the I data S 104 A and the Q data S 104 B. Here, the I data S 104 A and the Q data S 104 B maintain the time relationship so that their transition timings shift by a half cycle. Both data are NRZ signals having a signal rate that is half the spread data S 103 . The I-pulse shaping signal generator  105  and the Q-pulse shaping signal generator  106  generate the I-pulse shaping signal S 105  and the Q-pulse shaping signal S 106  as stepped triangle waves. The I-pulse shaping signal S 105  and the Q-pulse shaping signal S 106  are generated at the timings so that each peak value of the waveforms corresponds to the center of the I data S 104 A and the Q data S 104 B, respectively. 
   The seventh multiplier  131  multiplies the I data S 104 A and the cosine carrier S 107 A together to generate the BPSK modulated cosine carrier S 131 . The eighth multiplier  132  multiplies the Q data S 104 B and the sine carrier S 107 B together to generate the BPSK modulated sine carrier S 132 . The third multiplier  110  multiplies the BPSK modulated the cosine carrier S 131  and the I-pulse shaping signal S 105  together to output the BPSK modulated I pulse S 110 . The fourth multiplier  111  multiplies the BPSK modulated the sine carrier S 132  and the Q-pulse shaping signal S 106  together to output the BPSK demodulated Q pulse S 111 . The adder  112  uses polarities for subtracting the BPSK modulated Q pulse S 111  from the BPSK modulated I pulse S 110  to synthesize both pulses and generates the output signal S 112 . 
   The output signal S 112  generated from the transmitter  130  according to the third embodiment is essentially the same as that described in the first and second embodiments. That is to say, the output signal shows a constant envelope and can realize a transmitter with high power efficiency. No band-pass filters are needed for transmission. The transmission side is free from an inter-symbol interference. The signal rate of the I and Q pulses each is half the spread data. This doubles an interval of the I and Q pulses in themselves. The pulses are subject to little inter-symbol interference occurring in the band-pass filter at the reception side. 
   The following describes specific configuration examples of the I-pulse shaping signal generator  105  and the Q-pulse shaping signal generator  106  described in the first through third embodiments.  FIG. 8  shows the configuration of the pulse shaping signal generation circuit  200 .  FIG. 9  diagrams waveforms operating in the circuit  200 . 
   A pulse shaping signal generation circuit  200  in  FIG. 8  is used as the I-pulse shaping signal generator  105  and the Q-pulse shaping signal generator  106 . The pulse shaping signal generation circuit  200  has a first DFF (D flip-flop)  211 , a second DFF 212 , a third DFF 213 , a fourth DFF 214 , a first current source  201 , a second current source  202 , a third current source  203 , and a current-voltage conversion circuit  204 . 
   The first through fourth DFFs  211 ,  212 ,  213 ,  214  constitute a 4-stage Johnson counter  210  that operates synchronously with a reference clock S 101  ( FIG. 9(   a )) output from an oscillator  101 . As seen from waveforms of (b), (c), (d), and (e) in  FIG. 9 , Q outputs S 211 , S 212 , S 213 , and S 214  from the first through fourth DFFs of the Johnson counter  210  rise from low to high levels and fall from high to low levels in order in synchronization with the rise of the reference clock S 101 . 
   The Q outputs S 211 , S 212 , and S 213  from the first through third DFFs turn on or off output currents from the first through third current sources  201 ,  202 , and  203 . The current-voltage conversion circuit  204  adds output currents S 201 , S 202 , and S 203  to each for current-voltage conversion. This can generate a pulse shaping signal S 204  having a stepped triangle waveform as seen from  FIG. 9(   g ). Further, it is possible to change the waveform of the generated pulse shaping signal S 204  by weighting output currents from the first through third current sources  201 ,  202 , and  203 . 
   The pulse shaping signal generation circuit  200  shown in  FIGS. 8 and 9  exemplifies a specific configuration of the I-pulse shaping signal generator  105  and the Q-pulse shaping signal generator  106 . It is to be distinctly understood that the other configurations may be used to generate a similar waveform. 
   Further, the above-mentioned first through third embodiments have described the frequencies and cycles simply as examples. It should also be understood that the other values may be specified without departing from the spirit and scope of the invention. 
   The above-mentioned embodiments have described the configuration examples assuming the special communication apparatus for transmission or transmission and reception. Further, for example, a personal computer for various data processing may be mounted with a board or a card designed for the communication processing equivalent to the transmitter according to the embodiment. The computer may be provided with the software to perform the processing in the baseband section. 
   Since the present invention provides the constant envelope in an output signal, it becomes possible to use nonlinear amplifiers such as C-class amplifiers for transmission, realizing a transmitter with high power efficiency. 
   Since the narrow-band frequency spectrum is used, no band-pass filters are needed for transmission. Since waveforms are free from deterioration due to band-pass filters for transmission, it is possible to realize a UWB communication apparatus characterized by a high transmission rate. Since no band-pass filters are used, no transmission power loss occurs, making it possible to realize a transmitter with high power efficiency and low power consumption. 
   Further, it is possible to decrease effects of inter-symbol interference due to the use of non-Nyquist filters The UWB communication speed can be increased.

Technology Category: 5