Patent Document

FIELD OF THE INVENTION 
     The present invention relates in general to power supply circuits and components therefor, and is particularly directed to a diode emulator for a DC—DC converter in Discontinuous Conduction Mode (DCM). Prior to each PWM cycle, a tristate pulse (TriState, both switching devices are in the off state) is applied and the phase voltage polarity is sensed. The TriState pulse width is set by a closed loop circuit, and thereby incrementally adjusting the turn-off time of the commutating device according to the polarity of the phase voltage. As a result, it will effectively track the negative going, zero-crossing of the ripple current through the inductor and thereby effectively minimize loss of efficiency. 
     BACKGROUND OF THE INVENTION 
     FIG. 1 diagrammatically illustrates the general circuit configuration of a conventional DC—DC voltage buck converter as comprising a DC—DC controller  10 , which switchably controls the turn-on and turn-off of a pair of electronic power switching devices, respectively shown as an upper FET (P-MOSFET or N-MOSFET) device  20  and a lower FET (N-MOSFET) device  30 . These MOSFET switching devices have their drain-source paths coupled in series between first and second reference voltages (Vdd and ground (GND)). A common or phase voltage node  25  between the two power FETs  20 / 30  is coupled through an inductor  40  (which may typically comprise a transformer winding) to a capacitor  50  coupled to a reference voltage terminal (GND). The connection  45  between the inductor  40  and the capacitor  50  serves as an output node from which an output voltage Vout is derived. 
     The buck converter&#39;s DC—DC controller  10  includes a gate driver circuit  11 , that is operative to controllably turn the two switching devices  20  and  30  on and off, in accordance with a pulse width modulation (PWM) switching waveform (such as that shown at PWM in the timing diagram of FIG. 2) generated by a PWM logic circuit  12 . The upper FET device  20  is turned on and off by an upper gate switching signal UG applied by the gate driver  11  to the gate of the upper FET device  20 , and the lower FET device  30  is turned on and off by a lower gate switching signal LG applied by the gate driver  11  to the gate of the lower FET device  30 . 
     For the case of timing diagram of FIG. 2, the upper FET  20  is turned on in accordance with the rising edge of the PWM waveform and turned off in accordance with the falling edge of the PWM waveform, whereas the lower NFET  30  is turned on in accordance with the falling edge of the PWM waveform. During relatively light load conditions, where the ripple current IL through the inductor  40  is larger than the average inductor current, it is desired to revert to a basic DC—DC converter. This is effected by effectively replacing the lower switching FET  30  with a diode function—optimally turning off the lower switching device coincident with the negative-going zero-crossing of the inductor ripple current IL, so as to prevent current return flow back into the converter, and maximizing efficiency. 
     Prior art techniques to accomplish this diode transition operation may sense the ripple current flowing through the inductor  40  via node  45 , or may sense the phase voltage at node  25  and couple the sensed variation to a comparator. FIG. 1 shows the example where the phase node voltage Vp is coupled to a comparator  13 . Ideally, the comparator, which is enabled by the PWM logic circuit, will provide an output coincident with the negative-going, zero-crossing of the ripple current, in response to which the controller&#39;s output driver turns off the lower NFET switch. 
     Unfortunately, this technique is successful only at relatively low PWM frequencies, due to the propagation delay through the comparator. To obtain reasonably acceptable performance at relatively high PWM frequencies (e.g., on the order of 1 MHZ and above), it is necessary to use a comparator that requires a large current, which increases cost and is not practical for low power applications. 
     SUMMARY OF THE INVENTION 
     In accordance with the present invention, shortcomings of DC—DC buck converter diode emulators, including those described above, are effectively obviated by means of a variable current ramp-based diode emulator, that monitors the state of the phase voltage at the common node between the two switching devices and, incrementally with each PWM cycle, adjusts the turn-off time of the lower FET, until the monitored phase voltage indicates that the emulator is effectively tracking the negative going, zero-crossing of the ripple current through the inductor. 
     For this purpose, the diode emulator includes a phase voltage sample circuit that is coupled to receive a tristate pulse signal and a phase voltage signal. The tristate pulse signal starts just after turning-off the lower FET and prior to the front edge of the PWM pulse signal and terminates at the start of the PWM pulse. The phase voltage is derived from the common node between the two FETs. The rate at which the phase voltage Vp changes during the interval of the tristate pulse depends upon the properties of the FETs and inductor current. Due to the inherent body diode properties of the upper and lower FET switches and the properties of the inductor, the phase voltage will have a relatively positive edge at the tristate pulse if the lower FET is turned off too late. If the lower FET is turned off too early, however, the phase voltage remains low, being sensed as a second logical state. 
     The sensed phase voltage sample is coupled to the data input of a multibit up/down counter, which is sequentially clocked by the PWM signal. The up/down counter is used to control the rate of discharge of a lower power FET turn-off control capacitor, and thereby the time of occurrence of a turn-off signal for the lower FET, based upon whether the lower FET was turned-off too early or too late during the previous PWM cycle. 
     At each PWM pulse, the contents of up/down counter are either incremented or decremented, depending on the state of phase voltage as sampled/sensed by the TriState pulse. For a first binary state of the sensed phase voltage, indicating that in the previous cycle the lower power FET was turned off too late, the contents of the up/down counter will be ‘incremented’ by one bit at the next PWM pulse. For a second binary state of the sense phase voltage, indicating that in the previous cycle the lower power FET was turned off too early, the contents of the up/down counter are ‘decremented’ by one bit at the next PWM pulse. 
     The digital outputs of the up/down counter are coupled to relay drive inputs of relay coils of a set of relay switches. The switch contacts of the relay switches are coupled between to a charge/discharge node of the lower power FET turn-off control capacitor and outputs of a multiport current mirror. The current mirror has a further output coupled to the charge/discharge node of the lower power FET turn-off control capacitor, and is configured such that the currents at its output ports are binarily weighted in accordance with preselected weighting ratios relative to a reference input current. 
     This selective weighting of the mirror&#39;s output currents is defined in accordance with a prescribed capacitor discharge transfer function and serves to provide an adjustable (variable slope) ramp signal to a first input of a digital comparator. A second input of the digital comparator is coupled to receive the voltage VREF ( 308  in FIG.  3 ). The output of the digital comparator is coupled via a flip-flop to an output port, from which the lower FET turn-off signal is supplied to the controller. 
     In response to the tristate pulse signal, the phase voltage sample circuit senses the state of the phase voltage. As pointed out above, the phase voltage will produce a relatively high positive edge if the lower NFET is turned off too late, whereas if the lower NFET is turned off too early, the phase voltage will remain low. The sensed ‘digital’ state of the phase voltage is coupled to the up/down counter. At the next PWM pulse which begins at the termination of the tristate pulse, the contents of the up/down counter will be either incremented or decremented depending upon the state of the sensed phase voltage. 
     When a PWM pulse cycle begins, with the PWM being high, the lower power FET turn-off control capacitor will have been charged to a prescribed voltage. At the moment of the lower power FET turns on, the capacitor-charging switch is opened by the PWM pulse, so that the capacitor is no longer being charged. Also, the other relay switches are selectively closed in accordance with the contents of the up/down counter, so that the capacitor will discharge through one or more paths, as defined by the states of relay switches, with the magnitude of the resulting ramp current depending upon the binary weighting ratio of its associated current mirror output port of the multiport current mirror. 
     During its discharge into the current mirror, the lower power FET turn-off control capacitor will present to the first input of the digital comparator a voltage that decreases from an initial value and eventually drops below that applied to the second input of the digital comparator. When this happens, the output of the digital comparator change states, producing the lower power FET turn-off signal Toff that is supplied to the controller. For successive PWM cycles, as long as the value of phase voltage Vp is positive—indicating that the lower power FET was turned off too late in the previous cycle, the contents of the up/down counter will be continue to incremented. Then, in response to the sensed phase voltage no longer being positive—indicating that the lower power FET was turned off too early in the previous cycle, the up/down counter will be decremented. 
     Thus, the controlled, phase voltage-based incrementing and decrementing of the up/down counter presents a variable ramp current to the digital comparator, so that as the ramp current is varied over successive cycles of the PWM signal, it eventually attains a ‘dithered’ convergence of the lower power FET&#39;s turn-off signal Toff, that enables the emulator to very closely track the negative going, zero-crossing of the ripple current through the inductor, and thereby effectively minimize loss of efficiency of the converter. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 diagrammatically illustrates the general circuit configuration of a conventional DC—DC voltage buck converter; 
     FIG. 2 is a timing diagram associated with the operation of the DC—DC voltage buck converter of FIG. 1; 
     FIG. 3 is a logic—circuit diagram of the adjustable current ramp-based diode emulator in accordance with the present invention; and 
     FIG. 4 is a timing diagram associated with the operation of the diode emulator of FIG.  3 . 
    
    
     DETAILED DESCRIPTION 
     Attention is now directed to FIG. 3, which is a logic—circuit diagram of the adjustable current ramp-based diode emulator in accordance with the present invention. The diode emulator has a first input port  301  to which a DC input voltage Vin is coupled and a DC power terminal  305  to which a DC voltage Vss is applied. These DC voltages are distributed throughout the circuit for powering the various components of the emulator. An input port  302  is coupled to receive a tristate pulse signal (shown at TriState in the timing diagram of FIG.  4 ), which is generated just after turning-off the lower FET and prior to the start (rising edge) of the PWM pulse signal coupled to input port  306 , and terminates at the start of the PWM pulse. As described briefly above and as will be detailed below, the tristate pulse is employed to sample the phase voltage Vp, just after the lower FET turning-off. 
     The tristate pulse TriState and a further phase voltage Vphase (or Vp), which is coupled to input port  303 , are applied to respective inputs  321  and  322  of a NAND gate  320  within a phase voltage sample and hold circuit  310 . The phase voltage signal Vp is derived from the common node  25  between the two FETs  20 / 30  shown in FIG. 1. A prescribed logic state (e.g., the high state) of the tristate pulse TriState serves to enable the NAND gate  320 , so that it may sense the state of the Vp input  303 . 
     Due to the inherent (body diode) properties of the upper and lower FET switches and the properties of the inductor, the phase voltage Vp will produce a relatively positive edge, as shown at Vphase in the timing diagram of FIG. 4, if the lower power FET is turned off too late. This positive edge is sensed as a logical high or ‘1’ state. On the other hand, if the lower NFET is turned off too early, the phase voltage will remain low, being sensed as a logical low or ‘0’ state. 
     The phase voltage sample and hold circuit  310  is comprised of a cascaded connection of NAND gate  320  and a flip-flop  330  (comprised of cross-coupled NAND gates  340  and  350 ). NAND gate  320  has a third input  323  coupled to a first reset output  362  of a RESET circuit  360 . A second reset output  363  of RESET circuit  360  is coupled as a reset input to NAND gate  350  of flip-flop  330 . 
     The RESET circuit  360  is comprised of combinational logic circuitry including an inverter  365 , having an input coupled to the PWM input port  306 , and its output coupled to first reset output  362  and to a NAND gate  366 . The output of inverter  365  is further coupled through serial-coupled inverters  367  and  368  to NAND gate  366 . The output of NAND gate  366  provides the second reset output  363  that resets flip-flop  330 . Within the phase voltage sample and hold circuit  310 , NAND gate  320  is enabled on the falling edge of the PWM pulse, and disabled on its rising edge, while flip-flop  330  is reset on the falling edge of the PWM pulse. Flip-flop  330  has its Q output  331  coupled to a state (A) input  371  of a multibit up/down counter  370 . 
     The up/down counter  370  has its clock (CLK) input  372  coupled to the PWM input port  306 . In the non-limiting example of FIG. 2, up/down counter  370  is shown as comprising a three bit (eight state) counter having respective D2 (MSB), D1 and D0 (LSB) outputs  375 ,  376  and  377 . However, it should be observed that counter  370  is not limited to this or any particular code resolution. The contents of the up/down counter  370  are used to control the rate of discharge of a ‘lower power FET turn-off control’ capacitor  410 , and thereby the time of occurrence of a turn-off signal for the lower power FET, based upon whether the lower power NFET was turned-off too early or too late during the previous PWM cycle. 
     To this end, at each PWM pulse, the contents of up/down counter  370  are either incremented or decremented, depending on the state of input  371 , which represents the output of phase voltage sample and hold circuit  310 . For a first binary state of input  371  (e.g., a value of 1, as may be associated with a relatively high or positive value of the sensed phase voltage indicating that in the previous cycle the lower power NFET was turned off too late), the contents of up/down counter  370  will be ‘incremented’ by one bit at the next PWM pulse. For a second binary state of the up/down counter&#39;s input  371  (e.g., a value of 0, as may be associated with a negative value of sensed phase voltage—indicating that in the previous cycle the lower power NFET was turned off too early), the contents of the up/down counter  370  will be decremented by one bit at the next PWM pulse. 
     The respective D2, D1 and D0 outputs  375 ,  376  and  377  of up/down counter  370  are coupled to relay drive inputs  381 ,  391  and  401  of relay coils  382 ,  392  and  402  of a set of relay switches  380 ,  390  and  400 . Second ends of each of the relay coils are referenced to the voltage Vss applied to DC terminal  305 . Associated with the relay coils  382 ,  392  and  402  of relay switches  380 ,  390  and  400  are respective switch contacts  383 ,  393  and  403 , first ends of which are coupled to charge/discharge node  411  of capacitor  410 . 
     As will be described, for each cycle of the PWM pulse, the capacitor  410  is initially charged (via a switch  430 ) to a prescribed voltage (e.g., 1.5 VDC). Then, during the PWM pulse low state and with the lower power FET turned on, the capacitor  410  is selectively discharged through one or more paths including switches  380 ,  390  and  400 , in accordance with the states of the relay drive inputs  381 ,  391  and  401  of relay coils  382 ,  392  and  402  as defined by the respective D2, D1 and D0 outputs  375 ,  376  and  377  of the up/down counter  370 . 
     Switch contacts  383 ,  393  and  403  have second ends  384 ,  394  and  404  thereof respectively coupled through reverse blocking diodes  385 ,  395  and  405  to Vss and to respective output ports  421 ,  422  and  423  of a multiport current mirror  420 . Current mirror  420  has a fourth output  424  coupled to the charge/discharge node  411  of capacitor  410 , and a reference current input node  425  coupled to a reference current input port  307  to which a prescribed reference current Iref is supplied. Current mirror  420  is configured such that the output currents at its output ports  421 - 424  are binarily weighted in accordance with preselected weighting ratios relative to the input or reference current applied to port  307 . This selective weighting of the mirror&#39;s output currents is defined in accordance with a prescribed capacitor discharge transfer function for the capacitor  410 , and serves to provide a variable slope ramp signal which is coupled to a digital comparator  450 . 
     The charge/discharge node  411  of capacitor  410  is coupled to a first end  431  of a switchable contact  432  of relay switch  430 , and to a first (−) input  451  of digital comparator  450 . A second end  433  of switchable contact  432  is coupled to a charging voltage reference port  304  and through a resistor  455  to a second (+) input  452  of comparator  450 . It is also coupled to a reverse blocking diode  386  connected to Vss. The relay switch  430  has a relay coil  434  coupled between PWM port  306  and Vss port  305 . 
     The output  453  of the digital comparator  450  assumes a first binary state (e.g., logical ‘0’) as long as the voltage at its first (−) input  451  is greater the voltage at its second (+) input  452 . However, when the voltage at its first (−) input  451  is not greater than the voltage at its second (+) input  452 , the output  453  of digital comparator assumes a second binary state (e.g., logical ‘1’), which serves as an NFET turn-off control signal to the controller. Digital comparator  450  has its output  453  coupled to a D flip-flop  460 . The Q output  461  of flip-flop  460  is coupled to an output port  309  from which a turn-off signal Toff is supplied to the controller. Flip-flop  460  has its reset input  465  coupled to receive the PWM pulse supplied to PWM port  306 . 
     The diode emulator circuit of FIG. 3 operates as follows. In response to the TriState pulse, the signal TriState coupled to port  302  goes high, just (after turn off of the lower FET, and prior to turning-on the upper FET at the rising edge of the PWM pulse) applied to the PWM input port  306  (as shown in FIG.  4 ), NAND gate  320  is enabled, so that it may sense the state of the phase voltage node Vp input  303 . The rate at which the phase voltage Vp changes during the interval of the tristate pulse depends upon the properties of the power devices and is defined in accordance with the inductor current by dVp/dt=I/C. As discussed above, the phase voltage Vp will produce a relatively high positive edge, as shown at Vphase in the timing diagram of FIG. 4, if the lower NFET is turned off too late. This positive edge is sensed as a logical high or ‘1’ digital state of the phase voltage. If the lower NFET is turned off too early, the phase voltage will remain low, being sensed as a logical low or ‘0’ digital state of the phase voltage. This sensed Vp state is coupled to flip-flop  330  and applied from its Q output to the A input  371  of the up/down counter  370 . 
     At the next PWM pulse which, as shown in FIG. 4, begins at the termination of the tristate pulse Tristate, the contents of the up/down counter  370  will be either incremented or decremented depending upon the state of the A input  371  (the sensed phase voltage state). For a first binary state of input  371  (e.g., ‘1’ indicating that in the previous cycle, the lower power FET was turned off too late), the contents of up/down counter  370  will be incremented one bit by the PWM pulse. For a second binary state of up/down counter input  371  (e.g., ‘0’ indicating that in the previous cycle, the lower power FET was turned off too early), the contents of up/down counter  370  will be decremented one bit by the PWM pulse. The resultant count value as output by the respective D2, D1 and D0 outputs  375 ,  376  and  377  of the counter  370  now defines the states of the relay drive inputs  381 ,  391  and  401  of relay switches  380 ,  390  and  400 . 
     For the previous PWM pulse cycle, capacitor  410  will have been charged via switch  430  to a prescribed voltage (e.g., 1.5 VDC). When the lower power FET turns on, the capacitor-charging relay switch  430  is opened by the PWM pulse, so that the capacitor  410  is no longer being charged. Also, relay switches  480 ,  490  and  400  are selectively closed in accordance with the respective D2, D1 and D0 outputs  375 ,  376  and  377  of the up/down counter  370 . In addition to the current discharge path from node  411  to current mirror input  424 , capacitor  410  may discharge through one or more additional paths, as defined by the states of relay switches  380 ,  390  and  400 , and the magnitude of current through each path will depend upon the binary weighting ratio of its associated current mirror output port of current mirror  420 , as described above. 
     As capacitor  410  is discharged into current mirror port  424  and to any of the current mirror ports  421 - 423  through whichever one or more relay switches  380 ,  390  and  400  have been closed in accordance with the respective D2, D1 and D0 outputs  375 ,  376  and  377  of the up/down counter  370 , the voltage across capacitor  410  and applied to the first (−) input  451  of digital comparator  450  will decrease from its initial value (e.g., 1.5 VDC). Eventually, the voltage across capacitor  410  (and applied to comparator input  451 ) will drop below that applied through resistor  455  to the second (+) input  452  of comparator  450 . When this happens the output  453  of the digital comparator  450  changes state, and produces the lower power FET turn-off signal Toff that is supplied to the controller at output port  309 . 
     Thus for successive PWM cycles, as long as the value of phase voltage Vp is positive—indicating that the lower power FET was turned off too late in the previous cycle, the contents of the up/down counter  370  will be incremented (by one bit per PWM cycle). This continues until the value of the phase voltage is no longer detected as positive—indicating that the lower power FET was turned off too early in the previous cycle. At this point, the contents of up/down counter  370  will be decremented (by one bit). Should the next tristate pulse-based phase voltage measurement be positive, the contents of the up/down counter  370  will again be incremented by one bit, and so on. 
     It can be seen therefore, that this controlled, phase voltage-based incrementing and decrementing of the up/down counter  370  provides a variable ramp current input the comparator  450 . As the ramp current is varied over successive cycles of the PWM signal, it eventually produces a dithered convergence of the lower power FET turn-off signal Toff that enables the emulator to very closely track the negative going, zero-crossing of the ripple current through the inductor and thereby effectively minimize loss of efficiency of the converter. 
     While we have shown and described an embodiment in accordance with the present invention, it is to be understood that the same is not limited thereto but is susceptible to numerous changes and modifications as known to a person skilled in the art, and we therefore do not wish to be limited to the details shown and described herein, but intend to cover all such changes and modifications as are obvious to one of ordinary skill in the art.

Technology Category: 5