Patent Document

This is a continuation of U.S. patent application Ser. No. 10/886,771, filed Jul. 7, 2004, now U.S. Pat. No. 7,148,742, which is incorporated by reference herein in its entirety. 

   BACKGROUND OF THE INVENTION 
   This invention relates to circuitry that provides a reference voltage, and more particularly relates to reference voltage circuitry that is implemented in integrated circuitry. 
   In general, voltage detection circuitry prevents utilization circuitry (e.g., memory) from operating when the power supply voltage is too low for proper operation. At power-up (e.g., start up) the voltage detection circuitry suppresses operation of the utilization circuitry until the supply voltage reaches a predetermined voltage. When the supply voltage reaches the predetermined voltage, the voltage detection circuitry may enable the utilization circuitry by asserting, for example, a POWER-UP ENABLE signal. Likewise, during power-down, the voltage detect circuitry can de-assert the POWER-UP ENABLE signal when the supply voltage falls below the predetermined voltage, thereby disabling the utilization circuitry. 
   The point at which the voltage detection circuitry changes the state of the POWER-UP ENABLE signal is sometimes referred to as the trip-point. During power-up (sometimes referred to herein as ramp-up), the trip-point occurs when the supply voltage exceeds a predetermined voltage. Similarly, during power-down (sometimes referred to herein as ramp-down), the trip-point occurs when the supply voltage falls below the predetermined voltage. 
   In conventional voltage detection circuitry, such as that shown  FIG. 1 , the trip-point corresponds to when transistor  120  turns ON and OFF. For example, during power-up, the POWER-UP ENABLE signal goes HIGH when transistor  120  turns ON. POWER-UP ENABLE goes high because the voltage at the drain of transistor  120  (Node A) is pulled to ground when transistor  120  is turned ON. This LOW signal is then inverted by inverter  130  to provide a HIGH POWER-UP ENABLE signal. 
   A problem with circuitry  100  (of  FIG. 1 ) is that the point at which transistor  120  turns ON is subject to temperature and process variations. Process variation refers to the variance of one circuit to the next. For example, when transistors are fabricated, the threshold voltage may not be uniform for all transistors. One threshold voltage may be 0.7 volts, whereas another threshold voltage may be 0.8 volts. Other components, such as resistors, typically vary in resistance and equivalent series inductance. Thus, when circuitry  100  is constructed, the turn ON point of transistor  120  may vary from one circuit to the next. Moreover, changes in temperature can cause the trip-point of a particular circuit to vary.  FIG. 2  shows how trip-points can vary from one voltage detection circuit to another and how temperature changes can affect the trip point for a particular voltage detection circuit. 
   This gross variance in trip-points is undesirable and can potentially result in permanent damage to utilization circuitry. For example, if the trip-point occurs before the supply voltage reaches a predetermined voltage, this may force the utilization circuitry to draw excessive current to compensate for being enabled at too low a-voltage, potentially resulting in a circuit damaging current spike. 
   Therefore, it is an object of the invention to provide voltage detection circuitry that is insensitive to process and temperature variation. 
   It is also an object of the invention to provide voltage detection circuitry that has a substantially constant trip-point. 
   SUMMARY OF THE INVENTION 
   These and other objects of the invention are provided by voltage detection circuitry that utilizes bandgap circuitry to provide a substantially constant trip-point independent of temperature and process variations. 
   The bandgap circuitry according to the invention is an operational amplifier (OPAMP) based bandgap circuit capable of providing a substantially constant reference voltage for use in voltage detection circuitry. This substantially constant reference voltage is produced independent of temperature and process variations and provides an excellent foundation for providing a substantially constant trip-point. The trip-point may be attained by comparing the reference voltage to an input voltage derived, for example, from a voltage divider circuit. When the input voltage exceeds the stable reference voltage, the voltage detection circuitry may assert the POWER-UP ENABLE signal. The voltage detection circuitry may de-assert the POWER-UP ENABLE signal when the input signal falls below the stable reference voltage. 
   The stable reference voltage is generated by the OPAMP&#39;s bandgap voltage. As is known in the art, the bandgap voltage is derived from the difference in emitter voltage of two BJT transistors being used in combination with resistors and a differential amplifier. More specifically, one of the BJT transistors has a larger emitter area than the other BJT. When the same level of current is applied to both BJT transistors, the transistor with the smaller emitter area produces a larger base emitter voltage than the other transistor. This voltage difference is amplified and provided as the bandgap voltage. 
   In order for bandgap circuitry to generate the bandgap voltage, sufficient startup current is required to generate the voltage at the emitter of the BJT transistors. The requisite current needed to generate the bandgap voltage occurs when the supply voltage climbs to a predetermined voltage (e.g., 1.1 or 1.2 volts). Until the supply voltage reaches the predetermined voltage, the bandgap circuitry operates in its non-stable region and is unable to generate its bandgap voltage. However, when the supply voltage reaches the predetermined voltage, the bandgap circuitry operates at its stable operating region and provides the bandgap voltage as the reference voltage. 
   The circuitry of the present invention uses startup circuitry to provide current, in addition to the current provided by a source (e.g., V CC ) needed to activate the bandgap circuitry. That is, the startup circuitry enables the bandgap circuitry to operate in its non-stable operating region. To accomplish this, the startup circuitry selectively supplies current to one of the BJT transistors, the application of which causes the emitter voltage to rise. This increase in voltage activates a portion of the differential amplifier, which in turn activates a startup transistor that couples the bandgap circuit&#39;s reference node to the supply voltage (e.g., V CC ). 
   The bandgap circuitry initially clamps the reference node to the supply voltage until the supply voltage approaches the bandgap voltage, at which point, the bandgap circuitry ceases to follow the supply voltage and provides the bandgap voltage to the reference node. An advantage of the present invention is that the transition from the supply voltage to the bandgap voltage is smooth (i.e., voltage overshoot is minimal), whereas in prior art circuitry, the transition results in substantial overshoot. 
   The startup circuitry may be controlled by a FAST STARTUP signal generated by logic circuitry of the invention that determines when the state of the POWER-UP ENABLE signal changes. Specifically, the state of the FAST STARTUP signal may be the inverse of the POWER-UP ENABLE signal. Thus, when POWER-UP ENABLE is LOW, the FAST STARTUP signal is HIGH, which turns the startup circuitry ON. Similarly, when POWER-UP ENABLE goes HIGH, the fast startup circuitry is turned OFF. The ability to stop the supply of current being supplied to the aforementioned BJT transistor reduces current consumption (which is attributable to wasteful power consumption) and avoids potential interference with the operation of the bandgap circuitry when the supply voltage is HIGH. 
   The voltage detection circuitry can be used in power-down conditions, in addition to power-up conditions. During power-down, the voltage detection circuitry maintains the reference voltage at the bandgap voltage until the supply voltage drops below the voltage necessary for the bandgap circuitry to generate its bandgap voltage. When the supply voltage drops below the requisite voltage the startup circuitry may be activated so that the voltage detection circuitry tracks the supply voltage down to a power-off voltage. This way, the voltage detection circuitry provides some level of voltage as its reference voltage even when the supply voltage is too low to enable the bandgap circuitry to provide its bandgap voltage. 
   The voltage detection circuitry according to the invention may be adapted for use in systems that require power-up and/or power-down conditioning. For example, the voltage detection circuitry may used in connection with memory circuitry such as DRAM. 
   Further features of the invention, its nature and various advantages will be more apparent from the accompanying drawings on the following detailed descriptions. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a simplified schematic diagram of prior art voltage detection circuitry. 
       FIG. 2  shows a graph illustrating various trip-points obtained during the operation of voltage detection circuitry of  FIG. 1 . 
       FIG. 3  is a simplified schematic diagram of voltage detection circuitry according to the invention. 
       FIG. 4  is a graph showing the reference voltage being provided by the bandgap circuitry according to the invention. 
       FIG. 5  shows a graph illustrating the trip-point obtained during operation of the voltage detection circuitry of  FIG. 3  according to the invention. 
       FIGS. 6A and 6B  show a circuit diagram of the bandgap circuitry according to the invention. 
       FIG. 7  is a timing diagram of different modes of operation of the circuitry in  FIG. 6  according to the invention. 
       FIG. 8  is an illustrative block diagram of a system that incorporates the invention. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 3  shows a simplified schematic diagram of voltage detection circuitry  300  according to the invention. Circuitry  300  includes voltage divider  310  that provides a predetermined ratio of V CC as an input voltage to the negative terminal of comparator  330 . The predetermined ratio is defined by resistors  312 ,  314 , and  316 . As shown, the anode of resistor  312  is coupled to V CC and its cathode is coupled to the anode of resistor  316 . The anode of resistor  314  is coupled to the cathode of resistor  316  and its cathode is coupled to ground. The input voltage is provided at the connection (Node B) between resistors  312  and  316 . 
   Bandgap circuitry  320  is connected to the positive terminal (Node A) of comparator  330 . The output of comparator  330  is connected to inverter  332 . The output of inverter  332  (Node C) is connected to a first input of NAND gate  334  and to the gate of PMOS transistor  318 . The source of PMOS transistor  318  is coupled to the anode of resistor  316  and its drain is connected to the cathode of resistor  316 . Time out circuitry  340  is connected to a second input (Node D) of NAND gate  334 . The output of NAND gate  334  is connected to inverter  350 , which provides the POWER-UP ENABLE signal. 
   In general, bandgap circuitry  320  provides a reference voltage, V REF , to Node A so that comparator  330  can compare V REF  to the input voltage at Node B. The reference voltage provided by bandgap circuitry  320  may vary in voltage, initially following V CC , from a power-off voltage (e.g., about 0 volts) to the bandgap voltage V BG  (e.g., typically about 1.2 volts). The comparison performed by comparator circuitry determines whether the trip-point is triggered, at which point the state of POWER-UP ENABLE changes. For example, during ramp-up, the trip-point may occur when the voltage at Node B exceeds the voltage at Node A. During ramp-down, the trip-point may occur when the voltage at Node B equals or falls below the voltage at Node A. 
   The reference voltage being provided by bandgap circuitry  320  is now described in connection with  FIG. 4 , which is a graph of voltage versus time.  FIG. 4  shows a voltage waveform of V CC  that initially ramps up, remains constant, and then ramps down.  FIG. 4  also shows a voltage waveform of the reference voltage V REF , as provided by bandgap circuitry  320  in relation to V CC . 
   When V CC  is initially applied to circuitry  300 , there is a phase during which bandgap circuitry  320  operates in its non-stable region. During this phase, bandgap circuitry  320  provides V CC  as V REF  to comparator  330 . Bandgap circuitry  320  continues to supply V CC  as V REF  until sufficient voltage (e.g., 1.1 volts) enables bandgap circuitry  320  to transition to its stable operating region. Once in the stable operating region, bandgap circuitry  320  is able to generate a stable V BG , thereby holding V REF  to V BG . When V CC  ramps down, V REF  is held at V BG  until V CC  drops to a voltage that causes bandgap circuitry  320  to revert to its non-stable operating region. At this point of reversion, V REF  tracks V CC . 
   Note that throughout ramp-up and ramp-down of V CC , V REF  does not substantially exceed V BG . It is noted, however, that if V REF  does exceed V BG , its overshoot is negligible (i.e., on the order of 0.01 to 0.02 volts), as opposed to prior art bandgap circuits that overshoot V BG  by as much as one to three volts. An advantage realized by preventing the V REF  from substantially exceeding V BG  is that it provides a substantially constant trip-point, thereby preventing unnecessary delays in triggering the trip-point. A delay occurs, for example, when the desired trip-point is reached (i.e., V CC  has ramped up to a predetermined voltage), but V REF  continues to track V CC , thereby preventing comparator circuitry from registering the trip-point until V REF  is brought down to V BG . 
   A further advantage realized by tracking V CC  until it reaches V BG  is that it ensures that comparator  330  does not detect a voltage on Node B as being sufficient when in fact the voltage at Node B is actually insufficient to warrant driving POWER-UP ENABLE HIGH. This is achieved because the input voltage at Node B will be lower in voltage than V REF  because the input voltage is reduced by a predetermined ratio (e.g., 90 percent) as defined by voltage divider  310 . 
   The advantages realized by tracking V CC  while at the same time preventing V REF  from exceeding V BG  are provided by startup circuitry  370 . In general, startup circuitry  370  assists bandgap circuitry  320  by selectively supplying current to a BJT transistor being used to generate the bandgap voltage in connection with the operational amplifier. The application of this current assures that V REF  is coupled to V CC  substantially immediately at the start of power-up. 
   Referring back to  FIG. 3 , startup circuitry  370  may be coupled to bandgap circuitry  320  or may be included within bandgap circuitry  320 . Startup circuitry  370  receives FAST STARTUP signal  360  from Node E. It is noted that the state of FAST STARTUP signal  360  is the inverse of the state of POWER-UP ENABLE. During power-up, FAST STARTUP signal  360  is HIGH because the time out circuitry  340  forces the output of NAND gate  334  HIGH for a predetermined period of time. This HIGH signal activates startup circuitry  370 , thereby enabling it to assist bandgap circuitry  320 . A more detailed discussion of the operation of startup circuitry  370  is discussed below in connection with  FIG. 6 . 
   In operation, circuitry  300  has a substantially constant trip-point, such as that shown in  FIG. 5 .  FIG. 5  shows that the trip-point occurs when V CC  is about 1.45 volts to 1.47 volts. It is understood that the trip-point can be set using a variety of different techniques. For example, the trip-point can be adjusted by changing the resistance values of resistors  312  and  314 . The trip-point can be adjusted by inserting a gain stage between bandgap circuitry  320  and comparator circuitry  330 . For example, a capacitor divider gain stage or an OPAMP based divider can be used to decrease the reference voltage. A low voltage reference may be useful for low voltage operation memory chips. A gain stage coupled with an amplifier can be used to increase the reference voltage. 
   Referring to  FIGS. 3 ,  4 , and  5  the operation of circuitry  300  is now described. Initially, at power-up, the voltage at V CC  is at a power off voltage and POWER-UP ENABLE is LOW. In addition, time-out circuitry  340  provides a LOW signal to NAND gate  334  for a predetermined period of time (e.g., 5 μs). The LOW signal being applied to NAND gate  334  results in a HIGH output (because any LOW input to a NAND gate results in HIGH output). This HIGH signal is inverted by inverter  350  causing POWER-UP ENABLE to go LOW. Thus, POWER-UP ENABLE is prevented from going HIGH even if the voltage at Node B is sufficient to trigger the trip-point. Moreover, because the LOW signal is provided to NAND gate  334 , this results in a HIGH signal being applied to FAST STARTUP  360 . This HIGH signal activates startup circuitry  370 , thereby coupling V REF  to V CC  at the start of power-up. 
   When the predetermined time period expires, time-out circuitry  340  provides a HIGH signal to NAND gate  334  until it is reset. 
   Initially at power up, the output of comparator  330  is HIGH because the input voltage is less than V REF . This HIGH output signal is inverted by inverter  332 , providing a LOW signal at Node C. This LOW signal activates PMOS transistor  318 , shorting resistor  316 . When resistor  316  is shorted, the voltage provided on Node B is determined by V CC  and the resistance of resistors  312  and  314 . Note that in the case where resistor  316  is shorted, the voltage at Node B will be lower than the case where resistor  316  is not shorted. 
   As discussed above, startup circuitry  370  has clamped V REF  to V CC . Thus, V REF  follows V CC  until bandgap circuitry enters into its stable operating region and couples V REF  to V BG . The output of comparator  330  goes LOW when the voltage at Node B exceeds V REF . Note that the point in which the output of comparator  330  transitions from HIGH to LOW is a trip-point, as shown in  FIG. 5 . This LOW signal is inverted by inverter  332  and provided to NAND gate  334  as a HIGH input signal. In addition, this HIGH signal turns transistor  318  OFF. Assuming that time-out circuitry  340  has timed out, it provides a HIGH signal to the other input of NAND gate  340 . Application of two HIGH inputs to NAND gate  334  results in a LOW output signal that turns startup circuitry  370  OFF (because FAST STARTUP  360  is LOW). This LOW signal is inverted by inverter  350  to drive POWER-UP ENABLE HIGH. 
   During power down, the output of comparator  330  changes from LOW to HIGH when the voltage at Node B equals or falls below V REF . The HIGH signal is inverted by inverter  332 , pulling Node C LOW. The LOW voltage at Node C turns PMOS transistor  318  ON, shorting resistor  316 . The LOW voltage at Node C forces NAND gate  334  to output a HIGH signal that causes startup circuitry  370  to turn ON, pulling V REF  to V CC . Furthermore, the HIGH signal at the output of NAND gate  334  is inverted by inverter  350  to drive POWER-UP ENABLE LOW. 
   PMOS transistor  318  and resistor  316  are provided to take into account noise that may be present in V CC . Noise may cause V CC  to oscillate, potentially causing the output of comparator  330  to vacillate between HIGH and LOW. During startup, for example, transistor  318  is turned OFF and resistor  316  is no longer being shorted when POWER-UP ENABLE is asserted. This adds the resistance of resistor  316  to voltage divider  310 , thereby causing the voltage on Node B to increase proportional to the added resistance. Increasing the voltage on Node B helps prevent the noise on V CC  from inadvertently switching the output of comparator  330  back to HIGH, which would de-assert POWER-UP ENABLE. 
     FIGS. 6A and 6B  show a transistor diagram of bandgap circuitry  600  according to the invention. The large dashed lines shown in-both FIGS. indicate how the two FIGS. should be aligned with respect to each other. Thus, by aligning the dashed lines it is seen that  FIG. 6A  represents an upper portion of circuitry  600  and  FIG. 6B  represents a lower portion of circuitry  600 . Note that hereinafter  FIGS. 6A and 6B  are referred to collectively herein as  FIG. 6 . 
   General regions of circuitry  600  are delineated by dashed boxes  602 ,  640 , and  670  to facilitate the description of an embodiment of the invention. Box  602  generally refers to the operational amplifier portion of circuitry, box  640  generally refers to the driving portion of circuitry  600 , and box  670  generally refers to the current source portion of circuitry  600 . It is understood that the groupings are merely illustrative and are not limiting. For example, the startup circuitry according to the invention may be collectively represented by both boxes  640  and  670 . In another example, operational amplifier may include both boxes  602  and  640 . 
   The operational amplifier portion, as delineated by box  602 , includes PNP BJT transistors Q 1  and Q 2 , resistors R 1 , R 2 , and R 3 , and differential amplifier  610 . The operation of operational amplifiers is well known in the art and need not be discussed in detail here. The relevant operable portions of the operational amplifier will become apparent in the following discussion. However, it is now worth noting that Node A, which is formed between the emitter of transistor Q 1  and resistor R 1 , is coupled to SWITCH NODE # 3 . Furthermore, Node C, which is formed between the gates of PMOS transistors  611  and  612 , is coupled to the gate of PMOS transistor  676 . 
   Box  640 , which may form part of the startup circuitry according to the invention, includes circuitry that is operable to selectively couple V REF  to V CC . Box  640  includes PMOS transistor  642  having its source coupled to V CC , its gate coupled to Node D, which is formed between the drain of PMOS transistor  612  and the drain of NMOS transistor  613 , and its drain coupled to V REF . V REF  may be coupled to SWITCH NODE # 1 . 
   Box  670 , which may form part of startup circuitry according to the invention, includes circuitry that is operative to selectively provide current to the emitter of BJT transistor Q 1  (Node A). Box  670  includes PMOS transistor  672  having its drain coupled to SWITCH NODE # 2 , its source coupled to V CC , and its gate coupled to Node E, which is formed between the drain of PMOS transistor  676  and the drain of NMOS transistor  678 . 
   SWITCH NODE # 2  is shown in  FIG. 6  connected to SWITCH NODE # 3 , effectively coupling the drain of transistor  672  to Node A. It is noted that SWITCH NODE # 2  can be connected to SWITCH NODE # 1  to enable circuitry  600  to operate in a different mode. 
   The source of PMOS transistor  676  is coupled to V CC  and its drain is coupled to the drain of NMOS transistor  678 . The gate of NMOS transistor  678  is coupled to Node E and to the drain of NMOS transistor  684 . The source of NMOS transistor  678  is coupled to the gate and drain of NMOS transistor  680 . The source of NMOS transistor  680  is coupled to the source of PMOS transistor  685  and the source of NMOS transistor  684 . NMOS transistor  684  has its gate coupled to receive FAST STARTUP signal  360  (of  FIG. 3 ). The gate and drain of PMOS transistor  685  are connected to PNP BJT transistor Q 3 . 
   With reference to circuitry  600  of  FIG. 6  and the timing diagram of  FIG. 7  the operation of circuitry  600  will now be described. Before power-up, transistors  611 ,  612 ,  613 ,  614 , and  676  are biased to be turned OFF, the voltage at V REF  is equal to power off voltage (or ground), and the start-up current is nil. 
   At the start of power-up FAST STARTUP goes HIGH, causing transistor  684  to turn ON. When transistor  684  is ON, this short-circuits transistors  678  and  680 , thereby coupling Node E to transistor  685 . This coupling results in the application of a voltage, provided by the combination of transistors  685  and Q 3 , to Node E. When transistor  684  is ON, the voltage at Node E is kept at a voltage level that is sufficiently LOW enough to ensure that transistor  672  is ON, or at least partially ON. 
   When transistor  672  is ON (in this case partially ON), current from V CC  flows through transistor  672  to the emitter of transistor Q 1 . This current provides more startup current to transistor Q 1  than it would receive in the absence of startup circuitry  670  according the invention. This increase in current causes the voltage at Node A to rise, resulting in the activation of NMOS transistor  613 . When transistor  613  turns ON, the voltage at Node D is pulled to ground (at least temporarily), which causes transistor  642  (e.g., a startup transistor) to turn ON, thereby coupling V REF  to V CC . It is understood that the current being supplied to BJT transistor Q 1  occurs substantially immediately after power up is initiated, providing immediate V CC  tracking. 
   As V REF  is pulled up by V CC , the voltage at Nodes A and B are pulled up via resistors R 1  and R 2 , respectively. Initially, during startup, the voltage on Node A is higher than the voltage on Node B. This voltage differential causes the operational amplifier to operate in its non-stable region. The voltages on Nodes A and B equalize as V REF  is pulled to a voltage that approaches the bandgap voltage (e.g., about 1.2 volts) of circuitry  600 . Once Nodes A and B are equal or substantially equal, this causes the voltages at Nodes C and D to be equal or substantially equal, thereby turning transistors  611 ,  612 , and  676  ON. 
   When transistor  676  is turned ON, the voltage at the gate of transistor  672  (Node E) increases to nearly V CC , causing transistor  672  to be slightly turned off. As V CC  rises, the current being driven to the emitter of BJT transistor Q 1  decreases, thereby further assisting the operational amplifier in generating its bandgap voltage. Specifically, decreasing the current flow to BJT transistor Q 1  helps to further equalize the voltage on Nodes A and B because the additional startup current is no longer needed to activate circuitry  600 . 
   When the trip-point is triggered, FAST STARTUP goes LOW, thereby turning transistor  684  OFF. At this point, transistors  678  and  680 , which are no longer being short-circuited, assist in maintaining the voltage at Node E at V CC  to minimize start-up current. As a result, the voltage at Node E is driven to a voltage that results in turning transistor  672  completely OFF. Once OFF, current no longer flows through transistor  672  to the emitter of BJT transistor Q 1 , thereby reducing current consumption and avoiding potential interference with the operation of the bandgap circuitry. 
   It will be understood that the foregoing drain and source orientation and emitter and collector orientation of the transistors described herein is not intended to be limiting, but merely illustrative of one way such transistors can be constructed. Therefore, the terms “source,” “drain,” “emitter,” and “collector” are to be interpreted in their broadest sense. 
     FIG. 8  shows a system that incorporates the invention. System  800  includes a plurality of utilization circuitry  801  (e.g., DRAM), a processor  870 , a memory controller  872 , input devices  874 , output devices  876 , and optional storage devices  878 . Voltage detection circuitry according to the invention may be used, for example, to enable utilization circuitry  801 , processor  870 , or memory controller  872 . For example, DRAM circuitry (not shown) may be enabled by voltage detection circuitry according to the invention to prevent the DRAM circuitry from operating (e.g., perform read and write functions) if the supply voltage is below a predetermined voltage. Data and control signals are transferred between processor  870  and memory controller  872  via bus  871 . Similarly, data and control signals are transferred between memory controller  872  and utilization circuitry chips  801  via bus  873 . Input devices  874  can include, for example, a keyboard, a mouse, a touch-pad display screen, or any other appropriate device that allows a user to enter information into system  800 . Output devices  876  can include, for example, a video display unit, a printer, or any other appropriate device capable of providing output data to a user. Note that input devices  874  and output devices  876  can alternatively be a single input/output device. Storage devices  878  can include, for example, one or more disk or tape drives. 
   Thus, power-up detection circuitry that operates with a substantially constant trip-point is provided. One skilled in the art will appreciate that the present invention can be practiced by other than the described embodiments, which are presented for the purposes of illustration and not of limitation, and the present invention is limited only by the claims which follow.

Technology Category: 3