Patent Document

FIELD OF THE INVENTION 
     The present invention relates generally to integrated circuits. More particularly, it pertains to structures and methods for current sense amplifiers and current comparators with hysteresis. 
     BACKGROUND OF THE INVENTION 
     The use of voltage sense amplifiers with hysteresis for noise rejection is known. The simplest voltage sense amplifier is an operational amplifier in a positive feedback configuration. In the case of voltage hysteresis two different trip points (Tph and Tpl) are defined and circuits are designed such that when a high signal is to be recognized it must exhibit a voltage higher than Tph before it is recognized and declared a high signal. In a similar manner, before a low signal is recognized it must exhibit a low voltage lower than the second trip point Tpl. A simple illustration of this is provided in FIGS. 1A,  1 B, and  1 C. 
     In the quest for higher speed signaling it has recently been proposed to use current mode interconnections rather than voltage mode. The goal is to provide impedance matching on signal interconnection lines to reduce or avoid reflections and ringing on the lines. The technique proposed is matching termination of the signal line(s) to the signal receiver by using current mode interconnections and current mode sense amplifiers or current mode comparators. Signal interconnection and clock distribution lines with low controlled impedances are most amenable to current mode signaling. Metal lines separated from metal ground planes or metal power supply distribution planes (which are at AC ground) by oxide or other integrated circuit insulators will have low characteristic impedances of the order 50 or 75 ohms. To avoid reflections and ringing these need to be terminated in their characteristic impedance which requires sense amplifiers or receivers with low input impedances and implies small voltage swings on the lines. This is most easily accomplished by using current sense amplifiers which normally have a low input impedance. Rather than trying to sense the small voltage swings on the lines one can instead sense the current signal. Both single ended and differential configurations are possible. Current sense amplifiers have been described for use in SRAM&#39;s and in low impedance current-mode interconnections in CMOS integrated circuits with shielded interconnection lines. While this will reduce reflections and ringing it will not completely eliminate them. Also, this technique is still susceptible to noise transients. 
     For the reasons stated above, and for other reasons stated below which will become apparent to those skilled in the art upon reading and understanding the present specification, it is desirable to develop sense amplifiers or current comparators which are even less susceptible to induced noise, current reflections or ringing. 
     SUMMARY OF THE INVENTION 
     The above mentioned problems for high speed signaling as well as other problems are addressed by the present invention and will be understood by reading and studying the following specification. The present invention provides a current sense amplifier or current comparator with adjustable thresholds for the detection of valid signals coupled with the rejection of small noise current transients or reflections and ringing when using low impedance interconnections and/or current signaling. In particular, an illustrative embodiment of the present invention includes current sense amplifiers with hysteresis introduced as receivers for current mode signaling and/or clock distribution on low impedance integrated circuit interconnection lines. The introduction of hysteresis into the current sense amplifiers and/or receivers will allow them to discriminate against noise transients since the output will not change states unless the signal becomes more positive than a high trip point, Tph, or more negative than a low trip point, Tpl. 
     A first embodiment includes a current sense amplifier which has a first amplifier and a second amplifier. Each amplifier includes a first transistor of a first conductivity type and a second transistor of a second conductivity type, where the first and second transistors are coupled at a drain region. A signal input is coupled to a source region of the first transistor. A signal output node is coupled to the drain region of the first and the second transistor in the second amplifier. The signal output node is further coupled to a gate of a third transistor to introduce hysteresis for various values of an input current. 
     These and other method embodiments, aspects, advantages, and features of the present invention will be set forth in part in the description which follows, and in part will become apparent to those skilled in the art by reference to the following description of the invention and referenced drawings or by practice of the invention. The aspects, advantages, and features of the invention are realized and attained by means of the instrumentalities, procedures, and combinations particularly pointed out in the appended claims. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIGS. 1A,  1 B, and  1 C provide a prior art representation of high and low trip points for a voltage sense amplifier with hysteresis. 
     FIG. 2A is a schematic illustration of a conventional current sense amplifier. 
     FIG. 2B is a graphical representation of the current versus voltage (I-V) curve of the conventional current sense amplifier shown in FIG.  2 A. 
     FIG. 2C is another graphical representation of the current versus voltage (I-V) curve of the conventional current sense amplifier shown in FIG.  2 A. 
     FIG. 3A is a schematic illustration of a current sense amplifier, or current comparator, according to the teachings of the present invention. 
     FIG. 3B is an I-V graph illustrating one embodiment of the operation of the novel current sense amplifier circuit shown in FIG.  3 A. 
     FIG. 4A is a schematic illustration of another embodiment of a current sense amplifier, or current comparator, according to the teachings of the present invention. 
     FIG. 4B is an I-V graph illustrating one embodiment of the operation of the novel current sense amplifier circuit shown in FIG.  4 A. 
     FIG. 5A is a schematic illustration of another embodiment of a current sense amplifier, current comparator, or receiver with hysteresis provided for both negative and positive values of an input current I 1 . 
     FIG. 5B is an I-V graph illustrating one embodiment of the operation of the novel current sense amplifier circuit shown in FIG.  5 A. 
     FIG. 6 is a block diagram illustrating an electronic system according to the teachings of the present invention. 
     FIG. 7 illustrates, in flow diagram form, a method of forming a current sense amplifier according to the teachings of the present invention. 
     FIG. 8 illustrates, in flow diagram form, a method of forming a current comparator with hysteresis. 
     FIG. 9 illustrates, in flow diagram form, a method for operating a current sense amplifier according to the teachings of the present invention. 
     FIG. 10 illustrates, in flow diagram form, another method for operating a current sense amplifier according to the teachings of the present invention. 
    
    
     DETAILED DESCRIPTION 
     In the following detailed description of the invention, reference is made to the accompanying drawings which form a part hereof, and in which is shown, by way of illustration, specific embodiments in which the invention may be practiced. In the drawings, like numerals describe substantially similar components throughout the several views. These embodiments are described in sufficient detail to enable those skilled in the art to practice the invention. Other embodiments may be utilized and structural, logical, and electrical changes may be made without departing from the scope of the present invention. The following detailed description is, therefore, not to be taken in a limiting sense, and the scope of the present invention is defined only by the appended claims, along with the full scope of equivalents to which such claims are entitled. 
     FIG. 2A is a schematic illustration of a conventional current sense amplifier  200 . In FIG. 2A, the conventional current sense amplifier  200  is shown driven with a single ended or single sided input, I 1 . The other differential input, I 2 , is held at zero amperes. The output voltage (V 2 ) is given by −Zv(I 1 −I 2 ), where Zv is the transimpedance (Gain) for the conventional current sense amplifier  200 . This transimpedance, Zv, is very high until the output voltage, V 2 , clamps at either a high level or a low level. In operation, the conventional current sense amplifier  200  wants to be symmetrically balanced. A current, I 1 , injected into node  4  will see a high impedance looking into transistor T 5  and a lower impedance looking into transistor T 1 . Therefore, the injected current, I 1 , flows mostly into transistor T 1 . This will subtract, or reduce, the amount of current flowing down the left hand side of the conventional current sense amplifier  200  through transistor T 3 . In result, the potential at node  1  increases which places a higher potential on the gate of T 4 . As the gate potential of transistor T 4  increases, transistors T 2  and T 6  operate to pull the output voltage, V 2 , down toward ground. FIG. 2B is a graphical representation of the current versus voltage (I-V) curve of the conventional current sense amplifier  200  shown in FIG.  2 A. 
     FIG. 2C is another graphical representation of the current versus voltage (I-V) curve of the conventional current sense amplifier  200  shown in FIG.  2 A. In this embodiment, the conventional current sense amplifier is driven with two inputs, or a differential signal, where I 2 =−250 micro Amperes (μA). Here, the output voltage, V 2 , changes states when I 1 =−250 μA, so that (I 1 −I 2 ) first becomes positive and −Zv(I 1 −I 2 ) goes to the most negative value. 
     FIG. 3A is a schematic illustration of a current sense amplifier  300 , or current comparator  300 , according to the teachings of the present invention. As shown in FIG. 3A, the current sense amplifier  300  includes a first amplifier  310 , or left side  310 , and a second amplifier  320 , or right side  320 . Each amplifier,  310  and  320 , includes a first transistor of a first conductivity type, Mr. and M 2  respectively. Each amplifier,  310  and  320 , includes a second transistor of a second conductivity type, M 3  and M 4  respectively. In one embodiment, the first transistor of a first conductivity type, M 1  and M 2 , includes an n-channel metal oxide semiconductor (NMOS) transistor, M 1  and M 2 . In this embodiment, the second transistor of a second conductivity type, M 3  and M 4 , includes a p-channel metal oxide semiconductor (PMOS) transistors, M 3  and M 4 . Transistors M 1  and M 2  are driven by a gate potential at node  7 . Each amplifier,  310  and  320 , includes a current sink, shown in FIG. 3A as transistors M 5  and M 6  which are driven by a gate potential at node  6 . The first and second transistors, M 1  and M 3 , of the first amplifier  310  are coupled at a drain region,  321  and  322  respectively, to node  1 . 
     Node  1  couples the drain region,  321  and  322 , for the first and the second transistor, M 1  and M 3 , in the first amplifier  310  to gates,  340  and  341 , of the second transistor, M 3  and M 4 , in the first and the second amplifiers  310  and  320 . The first and second transistors, M 2  and M 4 , of the second amplifier  320  are coupled at a drain region,  323  and  324  respectively. In the embodiment shown in FIG. 3A, a signal output node  2  is coupled to the drain region,  323  and  324 , of the first and the second transistors, M 2  and M 4 , in the second amplifier  320 . In an alternative embodiment, the signal output node  2  can be coupled to the drain region,  321  and  322 , of the first and the second transistors, M 1  and M 3 , in the first amplifier  310 . As shown in FIG. 3A the signal output node is further coupled to a gate  380  of a third transistor M 8 . In one embodiment, the third transistor M 8  is an n-channel metal oxide semiconductor (NMOS) transistor M 8 . Each amplifier,  310  and  320 , includes a signal input node,  5  and  4  respectively, which is coupled to a source region,  325  and  326 , of the first transistors, M 1  and M 2 . 
     A source region,  327  and  328 , for the second transistors, M 3  and M 4  respectively, in the first and second amplifiers,  310  and  320 , is coupled to a voltage supply Vdd at node  3 . In one embodiment, a drain region  336  of the third transistor M 8  is coupled to a source region  328  of the second transistor M 4  in the second amplifier  320 . In this embodiment, a source region  337  of the third transistor M 8  is coupled to the signal input node  4  of the second amplifier  320 . In one embodiment, the signal input node  5  of the first amplifier  310  receives an input current, I 1 , and the signal input node  4  of the second amplifier  320  receives a reference current, I 2 . 
     FIG. 3B is an I-V graph illustrating one embodiment of the operation of the novel current sense amplifier circuit  300  shown in FIG.  3 A. The operation of the novel current sense amplifier circuit  300  is explained by reference to FIGS. 3A and 3B. The third transistor M 8  introduces a controlled hysteresis into the current sense amplifier  300  of FIG.  3 A. Beginning at the left hand side of the graph, FIG. 3B illustrates the output voltage, V 2 , at a high state, or first state, output voltage. The high, or first state, output voltage, V 2 , turns on third transistor M 8  which then drives an input current, IM 8 , into node  4 . In other words, the third transistor M 8  provides an input current, IM 8 , into node  4  which acts in conjunction with the reference current I 2 . The single ended input current, I 1 , must overcome this combination of the reference, or differential current, I 2 , and the input current, IM 8 , before the output voltage, V 2 , can change states. At this point, the switching action of the output voltage, V 2 , of the current sense amplifier  300  is given by V 2 =−Zv(I 1 −(I 2 +IM 8 )). The value of (I 1 −(I 2 +IM 8 )) must become non zero or positive for the output to switch, or go to the second state, e.g. low state. Due to the input current IM 8 , I 1  will not “trip” the state of the current sense amplifier  300  until I 1  exceeds a certain positive current value, i.e. a high trip point, shown at  350  in FIG.  3 B. As one of ordinary skill in the art will understand upon reading this disclosure, the size and doping levels of the third transistor M 8  can be varied to provide a set magnitude of input current, IM 8 , into node  4 . In this manner, the circuit design of the novel current sense amplifier  300  can be manipulated to introduce a range of hysteresis for positive input current, I 1 , values into the current sense amplifier  300 . The set hysteresis introduced, by the addition of the third transistor M 8 , allows the novel current sense amplifier  300  to discriminate against small transient noise values which would otherwise cause the current sense amplifier to switch states prematurely and provide an inaccurate output voltage, V 2 . 
     In reverse operation, the single ended input current, I 1 , is decreased from a higher positive value, e.g. above trip point value  350 . As shown in FIG. 3B, while the input current, I 1 , is above trip point  350  the output voltage, V 2 , will be at a low state, or second state, output voltage. In this low, second state, the voltage potential applied to gate  380  of the third transistor M 8  will not turn “on” transistor M 8 . Thus, the third transistor M 8  is effectively removed from the current sense amplifier circuit  300 . In the embodiment of FIG. 3A and 3B, node  4  will only see a reference current, I 2 , here held at zero amperes. In other words, the third transistor is not providing any input current, IM 8 , into node  4 . In reverse operation, the single ended input current, I 1 , must again upset the balance of the current sense amplifier  300 , but in the opposite direction, e.g. the input current, I 1 , must overcome the reference or differential current, I 2 , of zero amperes before the output voltage, V 2 , will again change states. At this point, the output voltage, V 2 , of the current sense amplifier  300  is given by V 2 =−Zv(I 1 −I 2 ). In this reverse direction, (I 1 −I 2 ) must become negative for the output voltage, V 2 , to switch back, or return to the high state, or first state, output voltage. I 1  will not “trip” the state of the current sense amplifier  300  until I 1  passes below a second current value, i.e. a low trip point, shown at  360  in FIG.  3 B. In the embodiment shown in FIGS. 3A and 3B, the output voltage, V 2 , will not change states until I 1  has reached zero. As one of ordinary skill in the art will understand upon reading this disclosure, the high and low trip points presented in connection with FIGS. 3A and 3B are given by way of illustration and not by way of limitation. Other high and low trip points can be achieved by varying the amount of hysteresis introduced by the third transistor M 8  and/or by varying the differential/reference signal I 2  of the novel current sense amplifier  300 . 
     FIG. 4A is a schematic illustration of another embodiment of a current sense amplifier  400 , or current comparator  400 , according to the teachings of the present invention. As shown in FIG. 4A, the current sense amplifier  400  includes a first amplifier  410 , or left side  410 , and a second amplifier  420 , or right side  420 . Each amplifier,  410  and  420 , includes a first transistor of a first conductivity type, M 1  and M 2  respectively. Each amplifier,  410  and  420 , includes a second transistor of a second conductivity type, M 3  and M 4  respectively. In one embodiment, the first transistor of a first conductivity type, M 1  and M 2 , includes an n-channel metal oxide semiconductor (NMOS) transistor, M 1  and M 2 . In this embodiment, the second transistor of a second conductivity type, M 3  and M 4 , includes a p-channel metal oxide semiconductor (PMOS) transistor, M 3  and M 4 . Transistors M 1  and M 2  are driven by a gate potential at node  7 . Each amplifier,  410  and  420 , includes a current sink, shown in FIG. 4A as transistors M 5  and M 6  which are driven by a gate potential at node  6 . The first and second transistors, M 1  and M 3 , of the first amplifier  410  are coupled at a drain region,  421  and  422  respectively, to node  1 . Node  1  couples the drain region,  421  and  422  for the first and the second transistors, M 1  and M 3 , in the first amplifier  410  to gates,  440  and  441  of the second transistors, M 3  and M 4 , in the first and the second amplifiers  410  and  420 . The first and second transistors, M 2  and M 4 , of the second amplifier  420  are coupled at a drain region,  423  and  424  respectively, and to a signal output node  2 . Each amplifier,  410  and  420 , includes a signal input node,  5  and  4  respectively, which is coupled to a source region,  425  and  426 , of the first transistors, M 1  and M 2 . In the embodiment shown in FIG. 4A, the signal output node  2  is coupled to the drain region,  423  and  424 , of the first and the second transistors, M 2  and M 4 , in the second amplifier  420 . As shown in FIG. 4A the signal output node is further coupled to a gate  430  of a third transistor M 7 . In one embodiment, the third transistor M 7  is a p-channel metal oxide semiconductor (PMOS) transistor M 7 . 
     A source region,  427  and  428 , for the second transistors, M 3  and M 4  respectively, in the first and second amplifiers,  410  and  420 , is coupled to a voltage supply Vdd at node  3 . In one embodiment, a source region  431  of the third transistor M 7  is coupled to a source region  427  of the second transistor M 3  in the first amplifier  410 . In this embodiment, a drain region  432  of the third transistor M 7  is coupled to the signal input node  5  of the first amplifier  410 . In one embodiment, the signal input node  5  of the first amplifier  410  receives an input current, I 1 , and the signal input node  4  of the second amplifier  420  receives a reference current, I 2 . 
     FIG. 4B is an I-V graph illustrating one embodiment of the operation of the novel current sense amplifier circuit  400  shown in FIG.  4 A. The operation of the novel current sense amplifier circuit  400  is explained by reference to FIGS. 4A and 4B. The third transistor M 7  introduces a controlled hysteresis into the current sense amplifier  400  of FIG.  4 A. Beginning at the right hand side of the graph, FIG. 4B illustrates the output voltage, V 2 , at a low state, or first state, output voltage. The low, or first state, output voltage, V 2 , turns on third transistor M 7  which then drives a current, IM 7 , into node  5 , the signal input node  5  for the first amplifier  410 . In other words, the third transistor M 7  provides an input current, IM 7 , into node  5 . A single ended input current, I 1 , injected into input signal node  5  is supplemented by the input current, IM 7 . In order for the current sense amplifier  400  to switch the state of output voltage, V 2 , the current injected into the signal input node  5  must upset, or “trip” the balance of the current sense amplifier  400 . In this embodiment, the signal input node  4  is held at a differential/reference signal, I 2 , of zero amperes. At this point, the output voltage, V 2 , of the current sense amplifier  400  is given by V 2 =−Zv((I 1 +IM 7 )−I 2 ). Here, the value of ((I 1 +IM 7 )−I 2 ) must become negative for the output voltage, V 2 , to go to a second state, or high state. 
     Because of the supplemented current, IM 7 , being driven by the third transistor M 7 , the input current I 1  will not “trip” the state of the current sense amplifier  400  until I 1  passes below a certain negative current value, i.e. a low trip point, shown at  460  in FIG.  4 B. As one of ordinary skill in the art will understand upon reading this disclosure, the size and doping levels of the third transistor M 7  can be varied to provide a set magnitude of input current, IM 7 , into node  4 . In this manner, the circuit design of the novel current sense amplifier  400  can be manipulated to introduce a range of hysteresis for negative values of input current I 1  into the current sense amplifier  400 . The set hysteresis introduced, by the addition of the third transistor M 7 , allows the novel current sense amplifier  400  to discriminate against small transient noise values which would otherwise cause the current sense amplifier to switch states prematurely and provide an inaccurate output voltage, V 2 . 
     In reverse operation, the single ended input current, I 1 , is increased from a lower value, e.g. below trip point value  450 . As shown in FIG. 4B, while the input current, I 1 , is below trip point  450  the output voltage, V 2 , will be at a high state, or second state, output voltage. In this high, second state, the voltage potential applied to gate  430  of the third transistor M 7  will not turn “on” transistor M 7 . Thus, the third transistor M 7  is effectively removed from the current sense amplifier circuit  400 . In the embodiment of FIG. 4A and 4B, node  4  will see a reference current, I 2 , here held at zero amperes. With the third transistor M 7  turned “off,” the third transistor M 7  is not providing any input current, IM 7 , into node  5 . As explained above, the single ended input current, I 1 , must upset the balance of the current sense amplifier  400  in the opposite direction in order for the current sense amplifier  400  to switch states again, e.g. the input current, I 1 , must overcome the differential signal, I 2 , of zero amperes. At this point, the output voltage, V 2 , of the current sense amplifier  300  is given by V 2 =−Zv(I 1 −I 2 ) since the third transistor M 7  is removed from the current sense amplifier circuit  400 . 
     In this reverse direction, (I 1 −I 2 ) must become positive for the output voltage, V 2 , to switch back, or return to the low state, or first state, output voltage. I 1  will not “trip” the state of the current sense amplifier  400  until I 1  passes above a certain current value, i.e. a high trip point, shown at  450  in FIG.  4 B. In the embodiment shown in FIGS. 4A and 4B, the output voltage, V 2 , will not change states until I 1  has reached approximately zero Amperes. As one of ordinary skill in the art will understand upon reading this disclosure, the high and low trip points presented in connection with FIGS. 4A and 4B are given by way of illustration and not by way of limitation. Other high and low trip points can be achieved by varying the amount of hysteresis introduced by the third transistor M 7  and/or by varying the differential/reference current signal I 2  of the novel current sense amplifier  400 . 
     FIG. 5A is a schematic illustration of another embodiment of a current sense amplifier  500 , current comparator  500 , or receiver  500  with hysteresis provided for both negative and positive values of an input current I 1 . As shown in FIG. 5A, the current sense amplifier  500  includes a first amplifier  510 , or left side  510 , and a second amplifier  520 , or right side  520 . Each amplifier,  510  and  520 , includes a first transistor of a first conductivity type, M 1  and M 2  respectively. Each amplifier,  510  and  520 , includes a second transistor of a second conductivity type, M 3  and M 4  respectively. In one embodiment, the first transistor of a first conductivity type, M 1  and M 2 , includes an n-channel metal oxide semiconductor (NMOS) transistor, M 1  and M 2 . In this embodiment, the second transistor of a second conductivity type, M 3  and M 4 , includes a p-channel metal oxide semiconductor (PMOS) transistor, M 3  and M 4 . Transistors M 1  and M 2  are driven by a gate potential at node  7 . Each amplifier,  510  and  520 , includes a current sink, shown in FIG. 5A as transistors M 5  and M 6  which are driven by a gate potential at node  6 . The first and second transistors, M 1  and M 3 , of the first amplifier  510  are coupled at a drain region,  521  and  522  respectively, to node  1 . 
     Node  1  couples the drain region,  521  and  522  for the first and the second transistors, M 1  and M 3 , in the first amplifier  510  to gates,  540  and  541  of the second transistors, M 3  and M 4 , in the first and the second amplifiers  510  and  520 . The first and second transistors, M 2  and M 4 , of the second amplifier  520  are coupled at a drain region,  523  and  524  respectively. In the embodiment shown in FIG. 5A, a signal output node  2  is coupled to the drain region,  523  and  524 , of the first and the second transistors, M 2  and M 4 , in the second amplifier  520 . As shown in FIG. 5A the signal output node is further coupled to a gate  530  of a third transistor M 7 . In one embodiment, the third transistor M 7  is a p-channel metal oxide semiconductor (PMOS) transistor M 7 . Each amplifier,  510  and  520 , also includes a signal input node,  5  and  4  respectively, which is coupled to a source region,  525  and  526 , of the first transistors, M 1  and M 2 . 
     A source region,  527  and  528 , for the second transistor, M 3  and M 4  respectively, in the first and second amplifiers,  510  and  520 , is coupled to a voltage supply Vdd at node  3 . In one embodiment, a source region  531  of the third transistor M 7  is coupled to a source region  527  of the second transistor M 3  in the first amplifier  510 . In this embodiment, a drain region  532  of the third transistor M 7  is coupled to the signal input node  5  of the first amplifier  510 . As shown in FIG. 5A, signal input node  5  of the first amplifier  510  receives an input current, I 1 , and the signal input node  4  of the second amplifier  520  receives a reference, or differential current signal, I 2 . 
     As shown in FIG. 5A the signal output node  2  is further coupled to a gate  580  of a fourth transistor M 8 . In one embodiment, the fourth transistor M 8  is an n-channel metal oxide semiconductor (NMOS) transistor M 8 . In one embodiment, a drain region  536  of the fourth transistor M 8  is coupled to a source region  528  of the second transistor M 4  in the second amplifier  520 . In this embodiment, a source region  537  of the fourth transistor M 8  is coupled to the signal input node  4  of the second amplifier  520 . 
     FIG. 5B is an I-V graph illustrating one embodiment of the operation of the novel current sense amplifier circuit  500  shown in FIG.  5 A. The operation of the novel current sense amplifier circuit  500  is explained by reference to FIGS. 5A and 5B. The third transistor M 7  and the fourth transistor M 8  introduce a controlled hysteresis into the current sense amplifier  500  of FIG.  5 A. Beginning at the right hand side of the graph, FIG. 5B illustrates the output voltage, V 2 , at a low state, or first state, output voltage. The low, or first state, output voltage, V 2 , turns on third transistor M 7  which then drives a current, IM 7 , into node  5 , the signal input node  5  for the first amplifier  510 . In other words, the third transistor M 7  provides an input current, IM 7 , into node  5 . A single ended input current, I 1 , injected into input signal node  5  is supplemented by the input current, IM 7 . In order for the current sense amplifier  500  to switch the state of output voltage, V 2 , the current injected into the signal input node  5  must upset, or “trip” the balance point of the current sense amplifier  500 . The signal input node  4  is held at a reference, or differential, current signal, I 2 , here zero amperes. At this point, the output voltage, V 2 , of the current sense amplifier  500  is given by V 2 =−Zv((I 1 +IM 7 )−I 2 ). The value of ((I 1 +IM 7 )−I 2 ) must become negative for the output voltage, V 2 , to go to a second state, or high state. Because of the supplemented current, IM 7 , being driven by the third transistor M 7 , the input current, I 1 , will not “trip” the state of the current sense amplifier  500  until I 1  passes below a certain negative current value, i.e. a low trip point, shown at  550  in FIG.  3 B. 
     As one of ordinary skill in the art will understood upon reading this disclosure, the size and doping levels of the third transistor M 7  can be varied to provide a set magnitude of input current, IM 7 , into node  5 . In this manner, the circuit design of the novel current sense amplifier  500  can be manipulated to introduce a range of hysteresis for negative values of input current I 1  into the current sense amplifier  500 . The set hysteresis introduced, by the addition of the third transistor M 7 , allows the novel current sense amplifier  500  to discriminate against small transient noise values which would otherwise cause the current sense amplifier to switch states prematurely and provide an inaccurate output voltage, V 2 . 
     In reverse operation, the fourth transistor M 8  acts to introduce a controlled hysteresis into the current sense amplifier  500  of FIG.  5 A. Begining at the left hand side of the graph, FIG. 5B illustrates the output voltage, V 2 , at a high state, or second state, output voltage. The high, or second state, output voltage, V 2 , turns on fourth transistor M 8  which then drives an input current, IM 8 , into node  4 . In other words, the fourth transistor M 8  provides an input current, IM 8 , into node  4  which acts in conjunction with the reference current signal I 2 . The single ended input current, I 1 , must overcome this combination of reference current signal, I 2 , and input current IM 8  before the output voltage, V 2 , can change states. At this point, the output voltage, V 2 , of the current sense amplifier  500  is given by V 2 =−Zv(I 1 −(I 2 +IM 8 )). The value of(I 1 −(I 2 +IM 8 )) must reach a positive sum for the output voltage, V 2 , to switch or return to the low state, or first state, output voltage. Due to input current IM 8 , input current, I 1 , will not “trip” the state of the current sense amplifier  500  until I 1  exceeds a certain positive current value, i.e. a high trip point, shown at  560  in FIG.  5 B. As one of ordinary skill in the art will understand upon reading this disclosure, the size and doping levels of the third transistor M 8  can be varied to provide a set magnitude of input current, IM 8 , into node  4 . In this manner, the circuit design of the novel current sense amplifier  500  can be manipulated to introduce a range of hysteresis into the current sense amplifier  500  for positive input current I 1  values. The set hysteresis introduced, by the addition of the third transistor M 8 , allows the novel current sense amplifier  500  to discriminate against small transient noise values which would otherwise cause the current sense amplifier to switch states prematurely and provide an inaccurate output voltage, V 2 . 
     As one of ordinary skill in the art will understand upon reading this disclosure, the high and low trip points presented in connection with FIGS. 5A and 5B are given by way of illustration and not by way of limitation. Other high and low trip points can be achieved by varying the amount of hysteresis introduced by third and/or fourth transistors, M 7  and M 8 , and/or by varying the differential/reference signal I 2  of the novel current sense amplifier  500 . FIG. 5A and 5B illustrate a novel current sense amplifier  500  with hysteresis for both negative and positive values of input current I 1  by the inclusion of both transistors M 7  and M 8 . By the use of fixed current values to drive either I 1  and/or I 2  and the addition of transistors M 7  and M 8  a wide variety of hysteresis conditions can be provided for signal detection. The high trip point Tph and low trip point Tpl can be set at either positive or negative current values. 
     FIG. 6 is a block diagram illustrating an electronic system  600  according to the teachings of the present invention. The electronic system  600  includes a processor, or processing unit  610  and a memory device  620 , e.g. a random access memory (RAM). A bus  630  communicatively couples the central processing unit  610  and the memory device  620 . In one embodiment, the bus  630  includes a system bus, a serial connection, or other bus. In one embodiment, the processor  610  and the memory device  620  are on a single semiconductor wafer. In an alternative embodiment, the processor  610  and the memory device  620  are on two separate semiconductor wafers. In one embodiment, the memory device  620  further includes a current sense amplifier, current comparator, or receiver circuit as described and presented in detail above in connection with FIG.  3 A. In an alternative embodiment, the memory device  620  further includes a current sense amplifier, current comparator, or receiver circuit as described and presented in detail above in connection with FIG.  4 A. In another alternative embodiment, the memory device  620  further includes a current sense amplifier, current comparator, or receiver circuit as described and presented in detail above in connection with FIG.  5 A. 
     FIG. 7 illustrates, in flow diagram form, a method of forming a current sense amplifier according to the teachings of the present invention. The method includes forming a first amplifier and a second amplifier electrically coupled together  710 . Forming each amplifier includes forming a first transistor of a first conductivity type and forming a second transistor of a second conductivity type. The first and second transistors are coupled at a drain region. Forming each amplifier includes forming a signal input coupled to a source region of the first transistor. The method further includes forming a signal output node coupled to the drain region of the first and the second transistors in the second amplifier where forming the signal output node includes coupling the signal output node to a gate of a third transistor  720 . In one embodiment, forming the signal output node further includes coupling the signal output node to a gate of a fourth transistor. In one embodiment, forming a first amplifier and a second amplifier electrically coupled together includes coupling the drain region for the first and the second transistors in the first amplifier to gates of the second transistor in the first and the second amplifiers. In one embodiment, coupling the signal output node to a gate of a third transistor includes coupling the signal output node to a gate of an n-channel metal oxide semiconductor (NMOS) transistor. 
     FIG. 8 illustrates, in flow diagram form, a method of forming a current comparator with hysteresis. The method includes forming a first amplifier and a second amplifier which are electrically coupled together  810 . Forming each amplifier includes forming a first NMOS transistor and forming a first PMOS transistor where the first NMOS transistor and the first PMOS transistor are coupled at a drain region. Forming each amplifier includes forming a signal input coupled to a source region of the first NMOS transistor in each amplifier. The method further includes forming a signal output node coupled to the drain region of the first NMOS transistor and the first PMOS transistor in the second amplifier where forming the signal output node includes coupling the signal output node to gates of a second NMOS transistor and a second PMOS transistor  820 . In one embodiment, forming a first amplifier and a second amplifier which are electrically coupled include coupling the drain region for the first NMOS and the first PMOS transistors in the first amplifier to gates of the first PMOS transistors in the first and the second amplifiers. 
     FIG. 9 illustrates, in flow diagram form, a method for operating a current sense amplifier according to the teachings of the present invention. The method includes providing a current signal to a first signal input of the current sense amplifier  910 . The method includes providing a reference signal to a second signal input of the current sense amplifier  920 . The method further includes providing a feedback from a signal output of the current sense amplifier to the second signal input such that providing a first feedback from the signal output to the second signal input introduces a hysteresis into the current sense amplifier in order to discriminate against noise transients  930 . In one embodiment, the method of FIG. 9 includes providing a second feedback from the signal output to the first signal input. In one embodiment, providing a second feedback from the signal output to the first signal input includes adjusting a low threshold voltage trip point (Tpl) in the current sense amplifier. In this embodiment, providing a first feedback from the signal output to the second signal input includes adjusting a high threshold voltage trip point (Tph) in the current sense amplifier. 
     FIG. 10 illustrates, in flow diagram form, another method for operating a current sense amplifier according to the teachings of the present invention. The method includes providing a current signal to a first signal input of the current sense amplifier  1010 . The method includes providing a reference signal to a second signal input of the current sense amplifier  1020 . The method further includes providing a feedback from a signal output of the current sense amplifier to the first signal input through a first transistor and to the second signal input through a second transistor such that providing a feedback from the signal output of the current sense amplifier to the first and the second signal inputs includes adjusting voltage thresholds for the detection of valid signals along with the rejection of small noise current transients or reflections and ringing in the current sense amplifier  1030 . In one embodiment, adjusting voltage thresholds for the detection of valid signals along with the rejection of small noise current transients or reflections and ringing includes adjusting a high threshold voltage trip point (Tph) in the current sense amplifier and includes adjusting a low threshold voltage trip point (Tpl) in the current sense amplifier. In one embodiment, providing a feedback from a signal output of the current sense amplifier to the first signal input through a first transistor includes adjusting a low threshold voltage trip point (Tpl) in the current sense amplifier. In this embodiment, providing a feedback from a signal output of the current sense amplifier to the second signal input through a second transistor includes adjusting a high threshold voltage trip point (Tph) in the current sense amplifier. 
     CONCLUSION 
     Thus, novel structures and methods for improving high speed signaling on and between integrated circuits has been described. The novel current sense amplifiers with hysteresis are fabricated according to a streamlined CMOS process technology. The introduction of hysteresis into the current sense amplifiers and/or receivers will allow them to discriminate against noise transients since the output will not change states unless the signal becomes more positive than a high trip point, Tph, or more negative than a low trip point, Tpl. 
     Although specific embodiments have been illustrated and described herein, it will be appreciated by those of ordinary skill in the art that any arrangement which is calculated to achieve the same purpose may be substituted for the specific embodiment shown. This application is intended to cover any adaptations or variations of the present invention. It is to be understood that the above description is intended to be illustrative, and not restrictive. Combinations of the above embodiments, and other embodiments will be apparent to those of skill in the art upon reviewing the above description. The scope of the invention includes any other applications in which the above structures and fabrication methods are used. The scope of the invention should be determined with reference to the appended claims, along with the full scope of equivalents to which such claims are entitled.

Technology Category: 3