Patent Document

CROSS REFERENCE TO RELATED DOCUMENTS 
     This application is related to patent application Ser. No. 09/515,286 by Tilley, et al., entitled “Method and Apparatus for Settling and Maintaining a DC Offset,” which is a continuation-in-part of pending application Ser. No. 09/290,564 filed Apr. 13, 1999, Tilley, et al., entitled “Method and Apparatus for Settling a DC Offset,” and also related to patent application Ser. No. 09/515,843 by Tilley, et al., entitled “Enhanced DC Offset Correction Through Bandwidth and Clock Speed Selection,” and Ser. No. 09/515,834 by Ferrer, et al., entitled “DC Offset Correction Adaptable to Multiple Requirements,” filed concurrently herewith, assigned to Motorola, Inc., and incorporated herein by reference. 
    
    
     FIELD OF THE INVENTION 
     This invention relates generally to radio communication receivers, and more specifically to a radio communication receiver incorporating DC offset correction loops for Zero IF or Direct Conversion architectures. 
     BACKGROUND 
     DC offset receivers and Zero IF receivers are discussed generally in U.S. Pat. No. 4,653,117 issued in March 1987, to Heck, U.S. Pat. No. 5,079,526 issued in January 1992, to Heck, et al.; in U.S. Pat. No. 5,483,691 issued in January 1996, to Heck, et al.; in U.S. Pat. No. 5,893,029 issued in April 1999 and in U.S. Pat. No. 6,006,079 to Jaffee et al. issued in December 1999, which are hereby incorporated by reference. 
     Direct Conversion Receivers (DCR) or Zero IF (ZIF) architectures function by mixing the desired RF or IF signal down to baseband, or some very low frequency offset from DC. Therefore, by definition for DCR or ZIF receivers, the mixer&#39;s Local Oscillator (LO) frequency is approximately equal to the desired input RF frequency. Thus, the magnitude of the baseband DC signal is proportional to that portion of the RF signal that is exactly equal to the LO frequency. Any variations in RF power due to environmental (fading, multi-path) or circuit functionality (AGC, mixer LO-to-RF isolation) will affect the Direct Current (DC) voltage level at baseband. It is important for the optimum performance of the receiver that variations in the baseband DC are compensated. This implies a need for a Direct Current Offset Correction Loop (DCOCL) which can unobtrusively compensate for any DC variations in the baseband signal path. 
     The baseband signal path may include a parallel I and Q channel configuration, where the Q signal is 90° out of phase with 1. Great effort is usually expended to maintain symmetry between the I and Q channel circuitry in order to minimize distortion products that result from amplitude or phase imbalances between the respective paths. This is especially true for analog modulation schemes, where distortion products cannot be eliminated through DSP arithmetic manipulation such as forward-error correction, or auto-correlation techniques. Each I and Q channel often includes a differential baseband path (I and {overscore (I)}, Q and {overscore (Q)}), where I/{overscore (I)} and Q/{overscore (Q)} maintain a 180° phase relationship to enhance common-mode noise immunity. Any DC offset between I and {overscore (I)}, or Q and {overscore (Q)} signals, is interpreted as a shift in the intrinsic DC for the I and Q channels respectively. 
     If the DC shift is severe enough, any demodulation technique (digital or analog) which requires an accurate reference for the I and Q signals will be degraded. Furthermore, severe DC shifts within the baseband path can impact circuit performance, degrading the selectivity of the baseband active filters, minimizing the AGC-free dynamic range, and even “railing” the I/Q signal against the baseband bias limits. This can be further complicated by the finite isolation between RF and Local Oscillator (LO) signals in the down mixers of many real world Zero IF (ZIF) or Direct Conversion Receivers (DCR). This finite isolation can cause the LO to mix with itself creating a baseband voltage proportional to the LO to RF isolation. This phenomenon has historically complicated ZIF design implementations since baseband DC offsets created in this manner vary with mixer performance over temperature, gain and LO drive level. 
     The continuously tracking closed loop offset correction strategies (which continuously track out DC variations) afford the advantage that all variations in DC voltages are continuously tracked out in real time, continuously “centering” the I/Q signal thus providing maximum AGC-free dynamic range of the baseband path. The disadvantage of a continuously tracking closed loop strategy is that it effectively introduces a High Pass Filter (HPF) response into the baseband filter response by tracking out all baseband signal variations below the DCOCL loop bandwidth. This high pass filter response induces an equivalent “notch” in the ZIF equivalent passband. The notch effectively “nulls out” FM Bessel components of the desired RF signal that are equal to the local oscillator frequency, thus causing distortion in the time domain demodulated signal. While these distortion products could easily be accommodated in digital modulation protocols, constant envelope analog modulation (FM, PM, etc) is seriously distorted by this “notch effect”. 
     U.S. Pat. No. 5,079,526 by Heck et al, attempts to address this condition by phase-locking the LO (the LO defines the notch location in the ZIF pass band) to the RF signal at a known LO-to-RF offset. The offset is selected to minimize any distortion products; however, the notch still exists and it becomes very cumbersome to effectively optimize the LO-to-RF offset for all operating environments. 
     U.S. Pat. No. 5,483,691 by Heck et al. optimizes AGC performance in a ZIF system to the exclusion of integrating the functionality into secondary loops unrelated to AGC. So, a need exists for an “optimized” digital DCOCL methodology integrating AGC and DC offset correction functionality that can be initiated on an “as required” basis. Such a method will provide elimination of a passband notch while maintaining an accurate baseband DC voltage. This in turn provides optimum performance for both analog FM and digital protocols. 
     Thus, there is a need for a baseband DC offset correction method which can be universally applied to ZIF and Direct Conversion receivers in a manner which is minimally disruptive of normal communications without the negative effects of the characteristic ZIF passband notch produced by continuously tracking out DC offset variations and without the RF to LO isolation issues of previous designs. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The features of the invention believed to be novel are set forth with particularity in the appended claims. The invention itself however, both as to organization and method of operation, together with objects and advantages thereof, may be best understood by reference to the following detailed description of the invention, which describes certain exemplary embodiments of the invention, taken in conjunction with the accompanying drawings in which: 
     FIG. 1 is a block diagram of a Zero IF receiver circuit in accordance with an embodiment of the present invention. 
     FIG. 2 is a block diagram of a system architecture including the Zero IF receiver integrated circuit, digital signal processor, and microprocessor controller in accordance with an embodiment of the present invention. 
     FIG. 3 is a flow diagram describing a method according to the present invention for conducting DC offset correction in accordance with one of two possible operational modes of the receiver of an embodiment of the present invention. 
     FIG. 4 shows a characteristic response for a prototype DCOCL and AGC for slotted protocol receiver operation. 
     FIG. 5 shows a characteristic response for DCOCL and AGC for non-slotted receiver operation. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     While this invention is susceptible of embodiment in many different forms, specific embodiments are shown in the drawings and will herein be described in detail, with the understanding that the present disclosure is to be considered as an example of the principles of the invention and not intended to limit the invention to the specific embodiments shown and described. In the description below, like reference numerals are used to describe the same, similar or corresponding parts in the several views of the drawings. 
     An embodiment of the present invention provides a method that is optimized to provide correction of any DC offset voltage errors in the baseband signal path based on the particular operating environment of a ZIF or DCR receiver. The operating environment of the receiver is partitioned into two main categories: deterministic slotted protocol operation and non-slotted random operation. For slotted system operation, the AGC and DCOCL are strategically configured to accommodate the particular protocol in which the receiver is operating with minimal disruption of normal communication. If the receiver is operating in a non-slotted environment where the reception of RF signals is random, then a different methodology is utilized that takes advantage of simultaneous AGC and DCOCL functionality. In each case, the offset is ultimately corrected in hardware by applying an appropriate compensating DC voltage to the signal path. Complementing this hardware methodology, a multiplicity of DSP algorithms are described that can be employed to compensate the I and Q to arithmetically equalize any residual offsets after execution of the hardware correction. This arithmetic correction takes place in the digital signal processing, for example, as used in decoding, digital filtering, etc. 
     The present invention, in one embodiment, uses a Sample and Hold DCOCL strategy which adopts an event-initiated correction with finite duration that is subsequently “fixed” until future corrections are initiated. Because this strategy uses one-time compensation, the difficulty with this strategy is in determining when to initiate the correction sequence. In general, the longer the time between the last baseband DC offset correction and reception of the received signal, the greater the probability that DC drift will have occurred, thus degrading receiver performance. Ideally, initiating a correction just before receiving a signal would ensure optimum performance. For slotted protocol applications (TDMA, FDMA, slotted PSK, etc) this is feasible, as the received signal is deterministic based on the communication protocol. But for classic two-way dispatch operation where message timing is completely random, it is impossible to know apriori when an incoming message is about to be received. 
     One method to address this problem is to initiate a correction sequence periodically, whether needed or not, to ensure proper operation. This brute-force approach will unnecessarily increase processing requirements for the receiver&#39;s controller(s) (e.g. microprocessor, DSP, etc.), which is undesirable. This is especially true as sophisticated multi-mode, multi-band radios are developed with High Speed Data (HSD) applications which intrinsically increase microcomputer processing demands. Furthermore, since AGC operation affects the DC offset error being corrected, it becomes desirable to integrate the functionality of the AGC into the offset correction sequence to ensure proper offset correction results. 
     Referring now to the drawings and specifically to FIG. 1, there is shown a simplified block diagram of a ZIF receiver system  100  utilizing an Automatic Gain Control (AGC) system and a DC Offset Correction Loop in accordance with one embodiment of the present invention. The AGC system of the present invention includes a set of adjustable gain baseband amplifiers  114  and  118 , an adjustable gain IF pre-amplifier  158 , and an AGC control circuit  122  for controlling the gain of the various amplifiers in the receive path (including RF, IF and baseband amplifier stages). 
     A first amplifier stage (generally a low noise amplifier), such as a gain adjustable IF pre-amplifier  158 , receives a radio frequency (RF) or intermediate frequency (IF) signal  102 , which it amplifies to produce signal  146 . This amplified signal at  146  is applied to down-mixers  106  and  108 . A phase-shifting circuit  104  receives a local oscillator (LO) signal  159  and produces an in-phase signal (1)  142  and a quadrature signal (Q)  144  (The (I)  142  and (Q)  144  signals being 90 degrees out of phase with respect to each other.). 
     The (I)  142  and (Q)  144  signals are applied to down mixers  106  and  108  for mixing with the input signal  146  from IF preamplifier  158 . Down mixers  106  and  108  then convert the signals from the RF or IF to baseband signals  148  and  150 . It should be noted that all RF or IF, LO and baseband signals may include differential signal pairs to provide maximum common mode noise rejection. For clarity, only a single signal path representing each of the differential signal pairs is shown. For example, baseband signal  148  and  150  may be composed of I and {overscore (I)}, Q and {overscore (Q)} respectively, where I and {overscore (I)} are 180° out of phase with each other, and {overscore (Q)} is 180° out of phase with signal Q. Lowpass filters  110 ,  112 ,  120 , and  124  filter the baseband signals  148  and  150  to remove interference and limit the noise bandwidth of the receiver  100 . Lowpass filters  110 ,  120 , and  112 ,  124  are distributed along the baseband I and Q channels respectively, and may be interactively programmable. Variable gain stages  114  and  118  separate each filter stage (additional gain stages and/or filtering stages may also be present, but are not shown), preferably with at least one of the baseband gain stage pair  114  and  118  being under gain control by a control circuit such as AGC control circuit  122 . The analog differential filtered signals  130  and  131  are sampled by the Analog-to-Digital Converter  190  for further processing and demodulation. I and Q data samples are then placed in a receive data register  192  to produce a stream of serial data output (generally to a digital signal processor). 
     The filtered baseband analog signals  130  and  131  are processed through an AGC detector (DET)  140  to provide control voltage  151 . The control voltage  151  is then used by the AGC control block  122  to generate operational dependent response voltages  152  and  156  to control the gain response of gain blocks (amplifiers)  114 ,  118  and  158 . The baseband gain of amplifiers  114 ,  118  and RF/IF preamplifier  158  can be independently adjusted based on the selected response programmed into AGC control block  122 . Baseband signals  130  and  131  (I and Q) also are processed by analog-to-digital converters  160  and  186  for subsequent use by the DC offset control blocks  162  and  168 . The DCOCL Control blocks  162  and  168  set voltages generated by the Operational Transconductance Amplifiers (OTA)  164  and  170  respectively. The output voltage of the OTAs set differential DC offset voltages of the baseband signals  148  and  150  to produce the minimum DC offsets of the filtered baseband signals  130  and  131 , respectively. The characteristic response of the DCOCL control blocks  162  and  168  can be selected by the radio&#39;s microprocessor or microcontroller (not shown) from a series of deterministic sequences including one-time single event corrections, or continuous “closed loop” corrections. 
     Since the control signals  130  and  131  are utilized by the AGC and DCOCL circuits, the AGC can be indirectly controlled by the DCOCL to provide for simultaneous operation of the AGC and DC offset correction for optimum operation during baseband correction for certain receiver operating environments. 
     The present invention can be utilized in either a slotted (deterministic) protocol communication environment or a non-slotted (non-deterministic) environment. In a slotted (deterministic) protocol, the receiver operation is synchronized with the received signal. In such a slotted system, there are generally time slots when either (1) no information is being transmitted to the receiver of interest, or (2) information being received by the receiver is irrelevant, or (3) information can otherwise be discarded without consequence or with minimal consequence. The present invention takes advantage of any such periods in the protocol to perform baseband DC offset correction in a minimally disruptive manner. An example of this type of communication is in TDMA (time division multiple access) communication systems. 
     In non-slotted (non-deterministic) systems, the receiver has no advance knowledge of when it might receive a transmission. An example of this type of system is a conventional AMPS (Advanced Mobile Phone Service) analog cellular telephone environment or a simplex two way radio environment wherein a communication can take place at any time. Since the baseband DC offset correction process disrupts the receiver&#39;s ability to receive incoming transmissions during the period of the Offset correction, different strategies are used for each communication environment. Embodiments of this invention effectively utilize AGC control and DCOCL control to coordinate the correction of the baseband DC offset in the radio receiver. 
     Referring now to FIG. 2, a simplified general system block diagram is shown illustrating the interface of receiver  100  to digital signal processing and microprocessor control stages. 
     FIG. 2 is a general system block diagram illustrating how receiver  100  may be interfaced to subsequent signal processing and control subsystems of the radio receiver. In the preferred implementation, an integrated Zero IF integrated circuit (ZIF IC) receiver subsystem  200  provides the functionality described in connection with FIG.  1 . Those of ordinary skill in the art will appreciate that although this invention is illustrated in the context of a Zero IF system, the invention is equally applicable to Direct Conversion Receivers. In this illustration, the duplicated circuitry used to separately process I and Q has been represented in simplified form to show the broad functionality. RF input signals are received by a low noise amplifier (LNA)  204  and passed to mixer  206  for down-conversion to baseband. The output  208  of mixer  206  includes the differential baseband I/Q signals. The I/Q signals are presented to the PMA (Post Mixer Amplifier)  210  (which performs the function of providing low noise gain to enhance baseband signal to noise ratio) prior to baseband low-pass filtering at filter  220 . The output of filter  220  is a filtered differential baseband I/Q signal having an intrinsic DC component. Analog signal  224  is formatted into an industry standard 3 line SSI (Synchronous Serial Interface) operating at a predetermined sampling rate (for example, 24 Ksps) at interface  226  for coupling to an external digital signal processor (DSP)  230 . The three line SSI signal includes clock, data and frame synchronization information representing the I and Q signals and AGC or other pertinent information. 
     DSP  230  can be any suitable commercially available, custom or semi-custom digital signal processor chip. DSP  230  functions to perform Quadrature demodulation, RSSI calculation, and DC averaging of the I and Q signals as will be discussed later. DSP  230  is interfaced to a controller  236  which may be a microcomputer, a microcontroller, ASIC or other suitable control processor. Controller  236  is used for a variety of purposes in the receiver. For purposes of this invention, it carries out the functions of initiating baseband DC offset correction sequences based upon the information received from DSP  230  via a parallel interface (e.g. eight bit parallel) therewith. 
     Controller  236  is coupled to the Zero IF IC  200  via a three line Serial Port Interface (SPI) bus  240  coupled to Serial Port Interface block  242 , in the preferred implementation. The SPI is typically a three line interface incorporating data, clock and chip enable signals to control both the DCOCL and the AGC. The information carried in the three line interface  240  is coupled through SPI  242  to a DCOCL control block  244  and an AGC control block  248 . AGC control  248  also receives a measure of the signal strength from “sum-of-squares” (SOS) detector  252  which monitors the signal at  224 . Similarly, an analog to digital converter (ADC)  256  supplies feedback from  224  to DCOCL control  244 . DCOCL control  244  provides output to an OTA (operational transconductance amplifier)  264  which provides a DC correction at the mixer output  208 . The AGC control  248  acts directly on the LNA  204  to adjust the gain/attenuation supplied by this amplifier. 
     It should be noted, at this point, that there are two control loops in action in this embodiment—the AGC control loop and the DC offset correction loop. Both loops derive input from the filtered differential baseband signal and the Controller  236  and apply correction at  204  and  208 , respectively. The DCOCL is thus nested within the AGC control loop and they are interdependent. That is, a change in the DC offset can affect the AGC and vice versa. 
     In operation, the DCOCL operates as follows. Under control of controller  236 , the DCOCL control  244  initiates a correction to the baseband DC offset when instructed by the controller  236 . This process may be initiated under a variety of circumstances; however, in the preferred embodiment, the process is initiated at two times. The first is when the receiver is first turned on. The second is whenever the DC offset exceeds a predetermined threshold. In the preferred embodiment, this threshold depends upon the correction resolution as set by the OTA  264 —that is, the resolution of DC output voltages which can be supplied by the OTA  264 . For example, a correction can be initiated whenever the baseband DC offset exceeds twice the DC equivalent value of the Least Significant Bit (LSB) of the ADC  256 . (Note that the resolution of output of the OTA  264  can be mapped to the resolution of ADC  256 .) It should be noted that the threshold setting of twice the DC equivalent value of the LSB of ADC  256  is suitable to the present implementations, but, other threshold settings may be equally suitable to other implementations. 
     When the baseband DC offset exceeds this threshold, the DCOCL control  244  is instructed by the controller  236  to initiate a baseband DC offset correction sequence. The offset correction sequence can utilize a binary search routine such as that described in U.S. patent application Ser. No. 09/515,286 to Tilley, et al., entitled Method and Apparatus for Settling and Maintaining a DC Offset, assigned to Motorola, Inc. and filed concurrently herewith, and to it&#39;s parent application Ser. No. 09/290,564 filed Apr. 13, 1999, entitled Method and Apparatus for Settling a DC Offset, assigned to Motorola, Inc., which are hereby incorporated herein by reference. As described in these references, the offset correction sequence could also be initiated by several possible factors, such as temperature, as well as the preferred change in DC offset. 
     Several possible averaging algorithms can be used by DSP  236  (or other devices in other architectures) to determine the current level of baseband DC offset correction and thereby determine whether or not to initiate a baseband DC offset correction sequence. These averaging algorithms fall into at least four groups and may depend upon the protocol being used according to the mode of operation of the receiver. The four groups are as follows: 
     1) Simple Integration. In this process, which is the system default, the I and Q signal values are sampled and their values are simply averaged over a predetermined period of time by dividing by the number of samples. 
     2) Envelope averaging. In this process, I and Q samples which fall within particular upper and lower value ranges are averaged independently over a specified period of time. This produces an upper limit average and a lower limit average. After this period of time, a cumulative average is calculated using the relationship: 
     
       
         Total Average=(Upper Limit Average+Lower Limit Average)/2. 
       
     
     This technique is particularly well suited for slow data rate digital signaling systems such as trunking control channels, Digital Private Line (DPL)™ (trademark of Motorola, Inc.) and other low speed data applications. 
     3) Slotted Time Averaging. In this technique, I and Q samples are taken from a finite number of N intervals of time in which the desired received signal is known to be absent. The I and Q data for each interval is then independently averaged to produce N interval averages. After this is done, the Total Average is computed as follows:          Total Average     =       ∑       X   n     _       N                            
     where={overscore (X n )} the I and Q averages for the interval n; and 
     where N=the number of intervals. 
     This technique is particularly well suited for TDMA protocols where eliminating the carrier for the averaging calculation is highly desirable because momentary increases in the received bit error rate (BER) may result when arithmetically tracking out the average I and Q values in DSP processes (such as demodulation, filtering, etc.). 
     4) Weighted Average—In this technique, individual I and Q samples may be weighted based upon a predetermined function. For example, greater weight can be given to samples closer to the previously calculated average. The weighted samples can then be averaged over a fixed period of time. 
     Those of ordinary skill in the art will appreciate that the above averaging techniques can be used independently or in any combination as might be advantageous to a particular system. Those of ordinary skill in the art will also appreciate that other averaging techniques can be devised without departing from the present invention. 
     Referring now to FIG. 3, a flow diagram illustrates the decision logic and sequence for determining which AGC and DC correction sequence is utilized and how each sequence is initiated according to the preferred implementation. The processing depicted in this diagram takes place in the controller  236  in this embodiment, but those of ordinary skill in this art will understand that this can be implemented using a variety of control configurations. 
     In FIG. 3, the process begins at  300  with the powering up of the receiver, which may generally form a part of a radio transceiver. At  304 , the ZIF IC  200  receives its initial programming via  240  from controller  236 . This initial programming initializes all of the ZIF IC subsystems for normal operation. At  308 , the ZIF IC is instructed to set the AGC to maximum attenuation. In the present embodiment, this can be accomplished in steps of 0.3 dB for the receiver front end, 3 dB for the baseband amplification and 15 dB for the IF section. However, the important thing is that any incoming signal be eliminated by producing adequate signal attenuation. 
     At  312 , the baseband DC offset correction sequence is initiated utilizing, for example, the binary search algorithm. At  318 , the I/Q DC average is computed at the DSP  230  (or, in the alternative, at controller  236 ) over a specified time period (for example, one second) and arithmetically compensates for any residual DC offset in I and Q data. This averaging also determines the initial DC reference for the I and Q baseband signals for use in block  330 . The I and Q average may also be used as a reference value in any arithmetic operations which need a DC reference (such as, for example, demodulation, filtering, frequency control, etc.). In the current architecture, such operations are carried out in the DSP  230 . 
     This computed average is held in the DSP  230 . The AGC is set to normal operation, and the signal attenuation is disabled at  322  completing the DC offset correction sequence. At  326 , the receiver operates in its standard mode of operation with the I and Q DC average being computed by the controller  236  (or DSP  230 ). The DSP&#39;s DC offset value is held constant at the computed value determined by the initial DC offset averaged in block  318 . This DC offset is held until a programming event determines that a new baseband DC offset correction sequence is to be initiated. In the preferred embodiment, this programming event is the determination that the baseband DC offset has drifted (e.g. due to temperature, oscillator drift, or other factors) so that the offset now exceeds a predetermined threshold value. This threshold can be, for example, twice the minimum resolution of ADC  256  or equivalently twice the least significant bit of ADC  256 . If this threshold is not exceeded at  330 , standard radio operation continues. If this threshold is exceeded at  330 , a determination is made as to the receiver&#39;s operational environment at  334 . 
     The first determination made in identifying the receiver&#39;s operational environment is determining whether the receiver is operating using a slotted or non-slotted protocol (i.e. a deterministic or non-deterministic protocol, respectively) at  334 . If the receiver is using a non-slotted protocol at  334 , the controller  236  waits until no on-channel carrier is present at  340 . This is done to minimize the chances that the DC offset correction sequence will disrupt receipt of an incoming message to the user. The controller  236  then presets the ZIF IC to engage a predetermined amount of artificial DC baseband offset. The amount required is simply enough to cause the AGC to fully engage a maximum amount of attenuation at  344 . In this scenario, the DC offset is artificially set and the AGC responds due to the nested loop nature of the architecture. The DC offset correction process then waits for a predetermined period of time (for example, 5 milliseconds) to permit the receiver to settle to a quiescent state. 
     At  348 , the controller  236  initiates the baseband DC offset correction sequence to correct the DC baseband offset in hardware (that is, by application of a compensating DC level by OTA  264 ) at  348  using, for example, the binary search algorithm. The process then waits (for example, for 5 milliseconds) for the DC offset correction and AGC to simultaneously settle out at  352 . The receiver now resumes l/Q data receipt from the ZIF IC  200  at  358  with data being supplied to the DSP  230  from interface  226 . The DSP computes the average I and Q values using an appropriate algorithm as described earlier over a predetermined time period, for example, one second. This new I/Q average DC value becomes the new reference value for use at block  330  and is stored at  362 . The new value is also used by DSP  230  for standard receiver operations at  326  such as demodulation, filtering, and other such arithmetic functions relying upon an accurate DC value. 
     If a slotted protocol is being used at  334 , the next determination made is a further categorization of the nature of the protocol. At  368 , it is determined whether or not the on-channel RF signal cycles on and off in this protocol (as, for example, in TDMA). If so, the system takes advantage of this to minimize or eliminate the likelihood of interfering with a communication by selecting a time when there is no on-channel RF to do the DC offset correction. If the on-channel RF cycles on and off at  368 , the system waits at  372  until the on-channel carrier is off. The off-channel RF input power is then measured at this time at  374 . This can be determined, for example, from the AGC circuit. Control then passes to  376 , where the AGC is set to attenuate the input RF power by an appropriate amount to eliminate the signal for practical purposes. For example, the AGC can be set to attenuate the RF by an amount equivalent to the RF input plus 10 dB (for example) so that the signal is eliminated for all practical purposes relating to this invention. Alternatively, the AGC could simply increase the attenuation by the maximum amount, however, this might increase the settling time when the AGC is reset. 
     Once the RF input is eliminated, control passes to  378  where a binary search is initiated to correct the baseband DC offset. The AGC is then reset (preferably using a fast reset algorithm) at  382  and control passes to  358 . After the process of  358  and  362 , standard receiver operation resumes at  326 . 
     If, at  368 , the on-channel RF does not cycle on and off, a slightly different approach is taken. The on-channel RF input power is measured (e.g. by the RSSI—Received Signal Strength Indicator, AGC or any other suitable technique for measuring signal strength) at  384 . The AGC attenuation is then set to eliminate the on-channel signal at  376  and the process proceeds as in the previous example. 
     FIG. 4 is a characteristic response of I and {overscore (I)} (I shifted by 180 degrees) signals undergoing a DCOCL DC voltage correction and the corresponding AGC response for a slotted protocol correction. The AGC and DCOCL corrections are independently controlled via the controller  236 . In this chart, the I and {overscore (I)} signals initially exhibit a distortion which is characteristic of DC offset. At time T 1 , the DC offset correction sequence begins wherein the AGC is set for maximum attenuation of the input signal and the binary search process begins. During the binary search (from time T 1  to T 2 ) the DC offset is seen to settle out so that I is approximately equal to {overscore (I)} is approximately equal to zero. At time T 3 , the AGC is set in a fast recovery mode to quickly move back to a setting at T 4  which permits an appropriate amount of gain in the amplifiers for normal operation, and the input signal is restored. The I and {overscore (I)} signals are now seen to be symmetrical about the zero axis since the DC correction is completed. 
     While this chart shows the effect on the DC offset correction of the I signals, it will be clearly understood that the baseband Q signals behave in an entirely similar way under this process. 
     Turning now to FIG. 5, the characteristic response of the I and {overscore (I)} signals undergoing a DCOCL DC voltage correction and the corresponding AGC response in a non-slotted protocol are illustrated. The AGC response is dependant (controlled by) the DCOCL response, which is selectable via the controller. In this example, the controller  236  sets the DCOCL to a specific (large) offset value at time T 5 . Prior to this time, I and {overscore (I)} are seen to again exhibit a DC offset, but the programmed value set by the controller  236  is rather large and results in an extreme divergence of the signals—perhaps railing the signals as shown. The AGC level is seen to rise indicating increasing attenuation between about T 5  and T 6 . At time T 7 , the DC offset correction process is initiated and carried out until approximately T 8 . During this time, the system quickly converges to a point where the DC offset is corrected, while simultaneously, the AGC self corrects due to the nesting of the correction loops as previously described. 
     Thus, the present invention provides a sample and hold type DC baseband offset correction method which can be universally applied to ZIF and Direct Conversion receivers which is minimally disruptive of normal communications. The baseband DC offset correction is accomplished without the negative effects of the characteristic passband notch produced by continuously tracking out DC offset variations. 
     Those of ordinary skill in the art will recognize that the present invention has been described in terms of exemplary embodiments based upon use of a programmed controller. However, the invention should not be so limited, since the present invention could be implemented using hardware component equivalents such as special purpose hardware and/or dedicated processors which are equivalents to the invention as described and claimed. Similarly, general purpose computers, microprocessor based computers, micro-controllers, optical computers, analog computers, dedicated processors and/or dedicated hard wired logic may be used to construct alternative equivalent embodiments of the present invention. Moreover, while a specific overall architecture has been disclosed, the present invention may be utilized on other architectures without departing from the present invention. 
     While the invention has been described in conjunction with specific embodiments, it is evident that many alternatives, modifications, permutations and variations will become apparent to those of ordinary skill in the art in light of the foregoing description. Accordingly, it is intended that the present invention embrace all such alternatives, modifications, equivalents and variations as fall within the scope of the appended claims.

Technology Category: 5