Patent Document

CROSS-REFERENCE TO RELATED APPLICATIONS 
   The present application is related to French Patent Application No. 04/01903, filed Feb. 25, 2004, entitled “POWER AMPLIFICATION CIRCUIT AND OPERATIONAL AMPLIFIER INCORPORATING SAID CIRCUIT”. French Patent Application No. 04/01903 is assigned to the assignee of the present application and is hereby incorporated by reference into the present disclosure as if fully set forth herein. The present application hereby claims priority under 35 U.S.C. §119(a) to French Patent Application No. 04/01903. 
   TECHNICAL FIELD OF THE INVENTION 
   The present invention relates to a power amplification circuit and an operational amplifier incorporating such a circuit. The power amplification circuit is used in the operational amplifier as power output stage, or output buffer. 
   BACKGROUND OF THE INVENTION 
   A high input impedance, a low output impedance and wide output current and output voltage dynamic ranges are some of the desired features for a power amplification circuit. 
     FIG. 1  is a circuit diagram of a differential operational amplifier known in the prior art. Such an operational amplifier comprises two cascaded amplification stages. An input stage, designated by GM, is formed by a differential transconductance amplifier  10 , receiving currents Ip and In, corresponding to input electric potentials Vp and Vn, on two positive and negative inputs, respectively. The transconductance amplifier  10  is connected between two voltage supply terminals, a positive supply terminal  5  and a negative supply terminal  6 , respectively, referenced with respect to an electric ground terminal M. Respective electric potentials V CC   +  and V CC   −  of the supply terminals  5  and  6  can, for example, be +2.5 V and −2.5 V. The currents Ip and In are of the order of a few microamps in absolute value. The input electric potentials Vp and Vn can vary between V CC   −  and V CC   + ; they form the input signals of the operational amplifier. 
   The output of the transconductance amplifier  10  forms the output of the input stage. It is connected to a node A that forms an input of the power output stage  100 . Capacitors  102  and  103  connect the node A to the supply terminals  5  and  6 , respectively. These stabilize the operation of the operational amplifier. 
   A node D forms an output of the power output stage  100  and is also an output of the operational amplifier. A load  101 , of value Z L , is connected between the node D and the ground terminal M. The load  101  is usually equivalent to a resistor connected in parallel with a capacitor (not shown) Z L  can, for example, have a modulus equal to 100 ohms. 
   The power amplification circuit that forms the power output stage  100  is designated in the following description as circuit  100 . It comprises two circuit modules  60  and  70 . The module  60  comprises two pnp-type bipolar transistors  61  and  62 , preferably identical to each other. The emitters of the transistors  61  and  62  are connected to the supply terminal  5  by identical resistors  63  and  64 , respectively, with a common value R. R can, for example, be equal to 1 kilo-ohm. The bases of the transistors  61  and  62  are connected to each other, and also to the collector of the transistor  61 . In other words, the transistor  61  is configured as a diode. The module  60  thus configured forms a well-known Widlar current source with outflowing currents. This operates as a current mirror: the currents flowing from the collectors of the transistors  61  and  62  are equal to each other. 
   The module  70  is also a Widlar current source, but with inflowing currents. It has a complementary structure to that of the module  60 . The module  70  thus comprises two npn-type bipolar transistors  71  and  72 , preferably identical to each other. Each of these transistors has an emitter connected to the supply terminal  6  by a resistor  73  and  74 , respectively. The resistors  73  and  74  have the same common value, which can also be the value R, but not necessarily. The respective bases of the transistors  71  and  72  are connected to each other and, in addition, to the collector of the transistor  71 . 
   A current source  7  is connected between the collectors of the transistors  61  and  71 . The positive terminal of the source  7  is connected to the collector of the transistor  71 , and the negative terminal of the source  7  is connected to the collector of the transistor  61 . The intensity I of the current delivered by the source  7  may, for example, be 200 microamps. 
   The circuit  100  also comprises a module  20  of the ‘push-pull’ type. The module  20  comprises two intermediate bipolar transistors  1  and  2 , pnp and npn respectively. The transistors  1  and  2  are preferably matched, in other words they have identical structures but have electrical doping types that are reversed with respect to each other. The bases of the transistors  1  and  2  are connected to each other and to the node A. The emitters of the transistors  1  and  2  are respectively connected to the collectors of the transistors  62  and  72 , at a node B and at a node C, respectively. Output transistors  3  and  4 , of npn and pnp type respectively, and preferably matched, have their bases connected, respectively, to the nodes B and C. The emitters of the transistors  3  and  4  are connected to each other and to the node D. The collectors of the transistors  3  and  4  are connected to the supply terminals  5  and  6 , respectively. 
   According to the known configuration of the circuit  100 , the collector of the intermediate transistor  1  is directly connected to the supply terminal  6 , and the collector of the intermediate transistor  2  is directly connected to the supply terminal  5 . 
   When the difference Vp−Vn between the input electric potentials Vp and Vn is positive and progressively increasing, the electric potential of the node A, denoted V A , is also positive and varies according to an amplification characteristic of the transconductance amplifier  10 . In practice, V A  is equal to a saturation value that depends on V CC   + . According to the known operation of the module  20 , the transistor  1  is then in an off state. According to the operation in current mirror mode of the module  60 , a current equal to I flows between the emitter and the collector of the transistor  62 . Consequently, a current I flows from the node B towards the base of the transistor  3 . Therefore:
 
 i 3=β3× I,   (1)
 
   where i 3  is the current flowing through the transistor  3  from the collector to the emitter of the latter, and where □ 3  is the current gain of the transistor  3 . 
   The output current of the circuit  100 , denoted i OUT , is then equal to i 3 −i 4 , where i 4  is the current flowing through the transistor  4 , from the emitter to the collector. The orientations of i 3  and i 4  are indicated in  FIG. 1 . i 3  and i 4  are positive. The value of i OUT  is limited by the value of i 3  given by equation (1). This value is frequently denoted by I SOURCE . It is reached when the input electric potential Vp is higher than the input electric potential Vn, and when the value of the impedance  101  is sufficiently low. 
   Symmetrically, when the input electric potential Vp is lower than the input electric potential Vn, the electric potential of the node A is negative. The current i OUT  is then negative and limited, in absolute value, by the value of i 4  given by the equation (2):
 
 i 4=β4× I,   (2)
 
   where β 4  is the current gain of the transistor  4 . This value is frequently denoted by I SINK . I SINK  thus defined is a positive value. 
   When the electric potential V A  reaches a sufficiently high value, the transistor  62  is in a saturated state. The maximum value that the electric potential V D  can reach at the node D is then V CC   + −V oh , with:
 
 V   oh   =U   BE (3)+ U   EC sat (62)+ R×I,   (3)
 
   where U BE ( 3 ) is the difference between the electric potentials of the base and of the emitter of the transistor  3 , and where U EC sat ( 62 ) is the difference between the electric potentials of the emitter and of the collector of the transistor  62  in the saturated state. V oh  is called the drop-out voltage and can reach 1 volt. In the following description, the potential V D  is called the output electric potential of the circuit  100 . 
   A drop-out voltage V ol , similar to V oh , limits the value that the potential V D  can take when Vp−Vn is negative. The minimum value of V D  is then V CC   − +V ol . The voltage V ol  obeys the expression:
 
 V   ol   =U   EB (4)+ U   CE sat (72)+ R×I,   (4)
 
   where U EB ( 4 ) is the difference between the electric potentials of the emitter and of the base of the transistor  4 , and where U CE sat ( 72 ) is the difference between the electric potentials of the collector and of the emitter of the transistor  72  in the saturated state. V ol  can also reach 1 volt. 
   Furthermore, the respective emitters and bases of the transistors  1  to  4  form a closed loop. The difference between the electric potentials V A  and V D  is therefore given by the following double equation:
 
 V   A   −V   D   =U   BE (1)+ U   BE (3)= U   BE (2)+ U   BE (4),  (5)
 
   where U BE (j) denotes the electric potential between the base and the emitter of the transistor j, for j=1, 2, 3 or 4. 
   In the idle state of the circuit  100 , in other words when Vp=Vn, no current flows out of the node D in the direction of the impedance  101  (i OUT =0), and the electric potentials of the nodes A and D are equal to each other. The currents flowing, respectively, between the node B and the base of the transistor  3 , and between the node C and the base of the transistor  4  are very low compared to the current I. A current i 1  equal to I therefore flows in the transistor  1 , from the emitter to the collector of the transistor  1 . Similarly, a current i 2  equal to I flows in the transistor  2 , from the collector to the emitter of the transistor  2 . It therefore follows from equation (5) that i 3 =i 4 =n×I, where n is the ratio of the respective emitter areas of the transistors  3  and  2 , or of the transistors  4  and  1 : 
   
     
       
         
           
             
               
                 n 
                 = 
                 
                   
                     
                       emitter 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       area 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       of 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       transistor 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       3 
                     
                     
                       emitter 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       area 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       of 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       transistor 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       2 
                     
                   
                   = 
                   
                     
                       emitter 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       area 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       of 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       transistor 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       4 
                     
                     
                       emitter 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       area 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       of 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       transistor 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       1 
                     
                   
                 
               
             
             
               
                 ( 
                 6 
                 ) 
               
             
           
         
       
     
   
   For example, n can be in the range  1  to  10 . 
   In the idle state of the circuit  100 , the total current drawn by the circuit  100 , denoted I CONS , is the current flowing between the power supply terminals  5  and  6 . It is equal to the sum of the currents flowing respectively in the resistors  63  and  64 , of i 2  and of i 3 . Therefore:
 
 I   CONS =(3+n) ×I.   (7)
 
   It is desirable to reduce this value of the total current drawn by a power amplification circuit in the idle state. 
   SUMMARY OF THE INVENTION 
   To address the above-discussed deficiencies of the prior art, one aim of the present invention thus consists in proposing a novel power amplification circuit that has a reduced current consumption in the idle state, for identical values of I SOURCE  and I SINK . 
   A subject of the invention is therefore a power amplification circuit that comprises:
         an input and an output;   a first circuit module comprising a first and a second transistor of a same first type and each having a first and a second main electrode and a control electrode, said first main electrodes of said first and second transistors being respectively connected to a first voltage supply terminal via first and second substantially identical resistors, the respective control electrodes of said first and second transistors being connected to each other, the control electrode of said first transistor being also connected to the second main electrode of said first transistor;   a second circuit module comprising a third and a fourth transistor of a same second type opposite to the first type and each having a first and a second main electrode and a control electrode, said first main electrodes of said third and fourth transistors being respectively connected to a second voltage supply terminal by third and fourth substantially identical resistors, the respective control electrodes of said third and fourth transistors being connected to each other, the control electrode of said third transistor also being connected to the second main electrode of said third transistor;   a current source connected to said second main electrodes of the first and third transistors;   a push-pull module comprising:   a first and a second intermediate transistor, respectively of the first and second types, each having a first and a second main electrode and a control electrode, the two control electrodes of the first and second intermediate transistors being connected to each other and to the input of the circuit, the first main electrodes of the first and second intermediate transistors being respectively connected to the respective second main electrodes of said second and fourth transistors, at a first and a second node, respectively;   a first and a second output transistor, respectively of the second and first types, each having a first and a second main electrode and a control electrode, the control electrodes of said first and second output transistors being respectively connected to said first and second nodes, said first main electrodes of the first and second output transistors being connected to each other and to the output of the circuit, said second main electrodes of the first and second output transistors being respectively connected to said first and second voltage supply terminals.       

   The power amplification circuit also comprises:
         a third circuit module having a first and a second input connected respectively to the second main electrode of said first intermediate transistor and to the first main electrode of said second transistor, and arranged such that an electric current flowing into said second input is identical to an electric current flowing into said first input, and   a fourth circuit module having a first and a second output respectively connected to the second main electrode of said second intermediate transistor and to the first main electrode of said fourth transistor, and arranged such that an electric current flowing from said second output is identical to an electric current flowing from said first output.       

   Thus, a part of the current flowing in the second resistor of the first circuit module is sent into said third circuit module operating in current mirror mode. The current that flows through the second transistor of the first circuit module is therefore reduced, as is the current flowing through said first intermediate transistor. This reduction leads to a reduction in the current flowing between the main electrodes of the first output transistor. This results in a reduction of the total current drawn by the circuit between the two voltage supply terminals. 
   In a symmetrical fashion, a part of the current flowing in said fourth resistor of the second circuit module originates from said fourth circuit module operating in current mirror mode. This results in reductions in the currents flowing between the main electrodes of said fourth transistor and of said second intermediate transistor, and consequently also a reduction in the current flowing between the main electrodes of the second output transistor. 
   Furthermore, the maximum absolute value of the output current of such a circuit is identical to that of a circuit according to  FIG. 1 . 
   In addition, such a circuit has drop-out voltages substantially equal to those of a circuit according to  FIG. 1 . 
   The dynamic range of the output current and of the output electric potential of the circuit are therefore not reduced. 
   Lastly, this circuit is especially simple and robust. It is therefore inexpensive to produce and has a long operational lifetime. 
   In the preferred embodiment, said third circuit module comprises a fifth and a sixth transistor of said second type, each having a first and a second main electrode and a control electrode. Said first main electrodes of said fifth and sixth transistors are respectively connected to the second voltage supply terminal via fifth and sixth substantially identical resistors. The respective control electrodes of said fifth and sixth transistors are connected to each other, and the control electrode of said fifth transistor is also connected to the second main electrode of said fifth transistor. Said second main electrodes of said fifth and sixth transistors respectively comprise said first and second inputs of said third circuit module. 
   Similarly, said fourth circuit module can comprise a seventh and an eighth transistor of said first type, each having a first and a second main electrode and a control electrode. Said first main electrodes of said seventh and eighth transistors are respectively connected to the first voltage supply terminal via seventh and eighth substantially identical resistors. The respective control electrodes of said seventh and eighth transistors are connected to each other, and the control electrode of said seventh transistor is also connected to the second main electrode of said seventh transistor. Said second main electrodes of said seventh and eighth transistors respectively comprise said first and second outputs of said fourth circuit module. 
   Advantageously, at least some of the transistors of the power amplification circuit are bipolar transistors. 
   Another subject of the invention is an operational amplifier comprising a power amplification circuit such as is described above, said circuit forming a power output stage of said operational amplifier. Such an operational amplifier has a reduced total current consumption and is capable of delivering an identical output current. 
   Before undertaking the DETAILED DESCRIPTION OF The INVENTION below, it may be advantageous to set forth definitions of certain words and phrases used throughout this patent document: the terms “include” and “comprise,” as well as derivatives thereof, mean inclusion without limitation; the term “or,” is inclusive, meaning and/or; and the phrases “associated with” and “associated therewith,” as well as derivatives thereof, may mean to include, be included within, interconnect with, contain, be contained within, connect to or with, interleave, juxtapose, be proximate to, be bound to or with, have, have a property of, or the like. Definitions for certain words and phrases are provided throughout this patent document,those of ordinary skill in the art should understand that in many, if not most instances, such definitions apply to prior, as well as future uses of such defined words and phrases. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     Other features and advantages of the present invention will become apparent in the following description of a non-limiting exemplary embodiment, making reference to the appended drawings, in which like reference numerals represent like parts, and in which: 
       FIG. 1 , already described above, is a circuit diagram of a known operational amplifier; 
       FIG. 2  is a circuit diagram of an operational amplifier comprising a power amplification circuit according to the invention; 
       FIGS. 3   a  and  3   b  are circuit diagrams of two circuit modules that can be employed in an operational amplifier according to  FIG. 2 ; 
       FIGS. 4   a  and  4   b  show variations in the maximum output current, in absolute value, for two circuits according, respectively, to  FIG. 1  and to  FIG. 2 ; and 
       FIG. 5  illustrates a preferred configuration of transistors that can be employed for a circuit module according to  FIG. 3   b.    
   

   Identical references used in several figures correspond to elements that are identical or that have analogous functions. 
   DETAILED DESCRIPTION OF THE INVENTION 
     FIGS. 2 through 5 , discussed below, and the various embodiments used to describe the principles of the present invention in this patent document are by way of illustration only and should not be construed in any way to limit the scope of the invention. Those skilled in the art will understand that the principles of the present invention may be implemented in any suitably arranged power amplification circuit or an operational amplifier incorporating such a circuit. 
   An operational amplifier according to  FIG. 2  comprises components that are identical to those in an operational amplifier according to  FIG. 1 , configured in the same manner. The description of these common components is not repeated here and reference will be made to the description of  FIG. 1  above. 
   According to the circuit diagram in  FIG. 2 , in the circuit  100  according to the invention, the connection between the collector of the intermediate transistor  1  and the voltage supply terminal  6 , together with the connection between the collector of the intermediate transistor  2  and the voltage supply terminal  5 , are eliminated. These are replaced by two additional circuit modules  80  and  90 . 
   The module  80  is connected to the voltage supply terminal  6  at a node  87 . It is recalled that the electric potential of the voltage supply terminal  6  is lower than that of the voltage supply terminal  5 . The module  80  has an input  85  connected to the collector of the intermediate transistor  1 , and an input  86  connected to the emitter of the transistor  62  at a node E. The module  80  is configured such that an electric current i 5  flowing between the node E and the input  86  is identical to an electric current i 1  flowing from the collector of the transistor  1 . In other words, i 5 =i 1  and the module  80  operates in current mirror mode. The flow direction conventions adopted for the currents i 1  and i 5  are indicated in  FIG. 2 . 
     FIG. 3   a  is a circuit diagram of circuit module that can be employed for the module  80 . This circuit module is of the Widlar current source type with inflowing currents. It comprises two npn-type bipolar transistors  81  and  82 . The emitters of the transistors  81  and  82  are connected to the voltage supply terminal  6  via substantially identical resistors  83  and  84 , respectively. The respective bases of the transistors  81  and  82  are connected to each other, and also to the collector of the transistor  81 . The transistors  81  and  82  are preferably identical to each other. The resistors  83  and  84  can have a value of 1 kilo-ohm, for example. 
   The module  90  is connected to the voltage supply terminal  5  at a node  97 . The module  90  has an output  95  connected to the collector of the intermediate transistor  2 , and an output  96  connected to the emitter of the transistor  72  at a node F. The module  90  is configured such that an electric current i 6  flowing between the output  96  and the node F is identical to an electric current i 2  flowing into the collector of the transistor  2 . In other words, i 6 =i 2  and the module  90  also operates in current mirror mode. The flow direction conventions adopted for the currents i 2  and i 6  are indicated in  FIG. 2 . 
     FIG. 3   b  is a circuit diagram of another circuit module that can be employed for the module  90 . This other circuit module is of the Widlar current source type with outflowing currents. The bipolar transistors  91  and  92  are of the pnp type and are preferably identical to each other. The emitters of the transistors  91  and  92  are respectively connected to the voltage supply terminal  5  via substantially identical resistors  93  and  94 . The respective bases of the transistors  91  and  92  are connected to each other, and also to the collector of the transistor  91 . The resistors  93  and  94  can also have a value of 1 kilo-ohm, for example. 
   The consequence of the addition of the nodes E and F in the modules  60  and  70 , respectively, is that these modules no longer operate as current mirrors. In the idle state of the circuit  100 , the current flowing between the node B and the base of the transistor  3  is negligible compared with the current flowing into the emitter of the transistor  1 . The current flowing in the resistor  64  is then 2×i 5 . The relationship between the voltages within the loop formed by the resistor  63 , the transistors  61  and  62 , and the resistor  64  is then:
 
 R×I+U   EB (61)= R× 2 ×i 5+ U   EB (62),  (8)
 
   where U EB ( 61 ) and U EB ( 62 ) represent the electric potential difference between the emitter and the base for the transistors  61  and  62 , respectively. Consequently: 
             i5   =       I   2     +       Δ   ⁢           ⁢     U   EB         2   ×   R           ,         where Δ U   EB   =U   EB (61)− U   EB (62).  (9) 
   Taking the numerical values of these variables leads to: 
   
     
       
         
           
             
               
                 
                   
                      
                     
                       
                         Δ 
                         ⁢ 
                         
                             
                         
                         ⁢ 
                         
                           U 
                           EB 
                         
                       
                       
                         2 
                         × 
                         R 
                       
                     
                      
                   
                   ⁢ 
                   
                     &lt;&lt; 
                     
                       I 
                       2 
                     
                   
                 
                 , 
               
             
             
               
                 ( 
                 10 
                 ) 
               
             
           
         
       
     
   
   where | |denotes the absolute value. Consequently, i 5  is approximately equal to I/2. From this, it follows that the current flowing through the transistor  62  in the idle state (i.e. when Vp=Vn) is also approximately equal to I/2. The current flowing through the transistor  62  is therefore approximately equal to half of the corresponding current in the case of a circuit according to  FIG. 1 . 
   An analogous reasoning to that which has just been presented, applied to the module  70  and to the transistor  2 , leads to i 2 =i 6 ≈I/2. 
   The total current I CONS  drawn by a circuit  100  according to  FIG. 2 , in the idle state (i.e. when Vp=Vn), between the voltage supply terminals  5  and  6 , is the sum of:
         the current flowing in the resistor  63 , in other words I,   the current flowing in the resistor  64 ,   the current flowing into the module  90  from the node  97 , and   the current flowing between the collector and the emitter of the transistor  3 .       

   According to the node E, the current flowing in the resistor  64  is approximately equal to 2×I/2, in other words I. According to the values determined above for the currents i 2  and i 6 , the current flowing into the module  90  from the node  97  is i 2 +i 6 ≈2×I/2≈I. Furthermore, according to the equation (5) still valid for a circuit  100  according to  FIG. 2 , and taking into account that in the idle state V A =V D , the result is that U BE ( 3 )=−U BE ( 1 ), and therefore that i 3 =n×I/2. Consequently:
 
 I   CONS =3 ×I +n×I/2=(3+n/2)× I.   (11)
 
   n again denotes the ratio of the emitter areas of the transistors  3  and  2 , which is equal to the ratio of the emitter areas of the transistors  4  and  1 . For a given value of n, the total current drawn by a circuit  100  according to  FIG. 2  is therefore lower by n×I/2 with respect to that of a circuit  100  according to  FIG. 1 , when a comparison is made between the equations (7) and (11). 
   Table 1 below indicates the values of I CONS  for different values of n, for two circuits according to  FIG. 1  and to  FIG. 2 , respectively. Table 1 also indicates the relative gain in total current consumption for a circuit according to  FIG. 2  relative to a circuit according to  FIG. 1 : 
   
     
       
             
             
             
             
           
         
             
               TABLE 1 
             
             
                 
             
             
               n 
               I CONS  (FIG.1) 
               I CONS  (FIG.2) 
               
                 
                   
                     
                       Gain 
                       = 
                       
                         
                           ΔI 
                           CONS 
                         
                         
                           
                             I 
                             CONS 
                           
                           ⁡ 
                           
                             ( 
                             
                               FIG 
                               . 
                               
                                   
                               
                               ⁢ 
                               1 
                             
                             ) 
                           
                         
                       
                     
                   
                 
               
             
             
                 
             
           
           
             
                1 
                4 × I 
               3.5 × I 
               13% 
             
             
                2 
                5 × I 
                 4 × I 
               20% 
             
             
                5 
                8 × I 
               5.5 × I 
               31% 
             
             
               10 
               13 × I 
                 8 × I 
               38% 
             
             
                 
             
           
        
       
     
   
   The expressions for I SOURCE  and for I SINK  given by the equations (1) and (2) are still valid for a circuit  100  according to  FIG. 2 . The maximum absolute value of the current i OUT  is therefore the same for two circuits  100  according to  FIGS. 1 and 2 , respectively. 
   Similarly, the equations (3) and (4) giving the expressions for the drop-out voltages V oh  and V ol  are still valid for a circuit  100  according to  FIG. 2 . 
   The characteristics of a circuit  100  according to  FIG. 2  are now compared with those of a circuit  100  according to  FIG. 1 . In order to carry out this comparison, the current I of the source  7  of the circuit  100  according to  FIG. 2  is adjusted such that the two circuits exhibit identical values of total current drawn I CONS . For this purpose, the current I for the circuit  100  according to  FIG. 2  is increased so as to compensate for the gain indicated in the last column of Table 1, for example for n=10. All the components present in each of the two circuits considered, which correspond to the same references in  FIGS. 1 and 2 , have identical values aside from the value I for the current source  7 . 
     FIG. 4   a  illustrates the variations of I SOURCE  as a function of the electric potential of the node D when the difference Vp−Vn is positive, for each of the two circuits considered. These are therefore output characteristic curves, in voltage-current coordinates, respectively for a circuit  100  according to  FIG. 1  and for a circuit  100  according to  FIG. 2 . Each of these characteristic curves is of the current-generator type that comprises a plateau up to a maximum value of the electric potential of the node D, fixed by the drop-out voltage V oh . The value of I SOURCE  in this plateau region is around 0.038 amps, for a circuit  100  according to  FIG. 1 , and around 0.051 amps for a circuit  100  according to  FIG. 2 . The increase in the value of I SOURCE  obtained is therefore around 34%. The slope of each curve in the plateau region is linked to a residual variation in the electric potentials of the electrodes of the transistors  1  and  62 . 
   These curves show, in addition, that the drop-out voltage V oh  has substantially the same value for the two circuits according to  FIG. 1  and to  FIG. 2 , respectively. Indeed, the plateaus corresponding to each of the two circuits have identical lengths. 
     FIG. 4   b  is analogous to  FIG. 4   a,  but when Vp−Vn is negative. It indicates the minimum values of the current i OUT  (these minimum values being negative) and therefore corresponds to −I SINK . 
   Preferably, when one of the circuit modules  80  or  90  is in the form of an integrated circuit on the surface of a substrate S, at least one of the transistors of this module has a vertical configuration. According to such a configuration, one main conduction direction of the transistor is substantially perpendicular to a surface of the substrate. 
     FIG. 5  shows a bipolar pnp-type transistor having a vertical configuration. The direction D, oriented towards the top of the figure, is substantially perpendicular to a surface S of a substrate  1000 . The substrate  1000  is of the p type. Several doped regions are layered within the substrate  1000  under the surface S, in the direction D. A p + -type region  1001  forms the emitter region of the transistor, an n type region  1003  forms the channel of the transistor, an intermediate region  1004  is of the p type, and a region  1007 , also of the p type and called the buried region, forms the collector of the transistor. A current flowing from the emitter to the collector of the transistor crosses the regions  1001 ,  1003 ,  1004  and  1007 : hence it flows parallel to the direction D. Lastly, a deep region  1010  is n-doped. 
   An emitter contact region E is disposed above the region  1001  and in electrical contact with it. 
   The region  1003  is contiguous with base contact regions B referenced  1002   a  and  1002   b.  The contact regions  1002   a  and  1002   b  are disposed at the surface S, on either side of the region  1003  and are n + -doped. 
   The region  1007  is connected to collector contact regions  1005   a  and  1005   b  via electrical connection regions  1006   a  and  1006   b.  The contact regions  1005   a  and  1005   b  are disposed at the surface S and are p + -doped. The electrical connection regions  1006   a  and  1006   b  are formed by p-wells. 
   Lastly, the deep region  1010  separates the transistor from the rest of the volume of the substrate  1000 . It is connected to contact regions  1008   a  and  1008   b,  disposed at the surface S, via electrical connection regions  1009   a  and  1009   b.  The contact regions  1008   a  and  1008   b  are n + -doped and the electrical connection regions  1009   a  and  1009   b  are formed by n-wells. The region  1010  can thus be biased to a predetermined electric potential. 
   Electrical isolation regions  1011 , for example made of silica, separate the various contact regions at the surface S. 
   A transistor with such a configuration occupies a reduced area of the surface S of the substrate  1000 : it can be integrated to an especially high level. One dimension of the transistor taken parallel to the surface S of the substrate  1000  can be, for example, around 0.25 microns, as measured between the outer edges of two isolation regions  1011  at opposing ends of the transistor. The circuit module comprising such a transistor is therefore especially compact. 
   The configuration of a transistor illustrated by  FIG. 5  is presented by way of an example. Other equivalent configurations, that afford equivalent possibilities for integration, can be employed. 
   The circuit modules corresponding to  FIGS. 3   a  and  3   b  are also presented by way of examples. It will be understood that other circuits operating in current mirror mode and known to those skilled in the art can be used for the modules  80  and  90 . 
   Finally, it will also be understood that a circuit according to the principle of the invention can be obtained by employing field effect transistors, notably those using MOS (Metal Oxide Semiconductor) technology. The substitution of each bipolar transistor described above by a field effect transistor can be effected according to the known rules of correspondence between the various types of bipolar transistor and the various types of field effect transistor. It is intended that the present invention encompass such changes and modifications as fall within the scope of the appended claims.

Technology Category: 5