Patent Document

This application is a continuation-in-part which discloses and claims subject matter disclosed in my earlier pending application Ser. No. 09/312,091 filed May 15, 1999, now U.S. Pat. No. 6,147,886. This invention was revealed in Disclosure Document Nr. 460,696 filed Aug. 16, 1999. 
    
    
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The subject invention generally pertains to electronic power conversion circuits, and more specifically to high frequency, switched mode power electronic converter circuits. 
     2. Description of Related Art 
     There are some power conversion circuits which accomplish higher efficiencies by implementing a mechanism that accomplishes switching at zero voltage. Power loss in a switch is the product of the voltage applied across the switch and the current flowing through the switch. In a switching power converter, when the switch is in the on state, the voltage across the switch is zero, so the power loss is zero. When the switch is in the off state, the power loss is zero, because the current through the switch is zero. During the transition from on to off, and vice versa, power losses can occur, if there is no mechanism to switch at zero voltage or zero current. During the switching transitions, energy losses will occur if there is simultaneously (1) non-zero voltage applied across the switch and (2) non-zero current flowing through the switch. The power losses associated with the switching transitions will be the product of the energy lost per transition and the switching frequency. The power losses that occur because of these transitions are referred to as switching losses by those people who are skilled in the art of switching power converter design. In zero voltage switching converters the zero voltage turn off transition is accomplished by turning off a switch in parallel with a capacitor and a diode when the capacitor&#39;s voltage is zero. The capacitor maintains the applied voltage at zero across the switch as the current through the switch falls to zero. In the zero voltage transition the current in the switch is transferred to the parallel capacitor as the switch turns off. 
     The zero voltage turn on transition is accomplished by discharging the parallel capacitor using the energy stored in a magnetic circuit element, such as an inductor, and turning on the switch after the parallel diode has begun to conduct. During the turn on transition the voltage across the switch is held at zero clamped by the parallel diode. The various zero voltage switching (ZVS) techniques differ in the control and modulation schemes used to accomplish regulation and in the energy storage mechanisms used to accomplish the zero voltage turn on transition. 
     One of the ZVS techniques uses a resonant circuit which is frequency modulated over a broad frequency range. An example is shown in FIG.  1 . These techniques have been refined by a multi-resonant technique in which more resonant circuit elements and a complex control circuit are required, but the converter can operate at a fixed frequency. 
     Several techniques have been introduced which accomplish zero voltage switching, inherently, at constant switching frequency. One of these techniques requires a full bridge switching arrangement with four primary switches in which the regulation is accomplished by phase modulation or by alternating pulse width modulation in the two switching legs. This technique is illustrated in FIG.  2 . This technique has become a standard technique for high power conversion at high frequency. One of the potential problems with this technique is staircase saturation of the transformer core resulting from relatively small DC imbalances in the transformer&#39;s primary winding, which can lead to catastrophic failure. One common solution to the staircase saturation problem is to place a capacitor in series with the primary winding of the transformer to block any DC current. The series capacitor incurs losses during high power operation and requires the user to use voltage mode control rather than the preferred current mode control, which unbalances the capacitor voltage resulting in high switch stress that can lead to catastrophic failure. Another problem is high conduction losses that results from the peak recirculation current during the reset time of the output choke. 
     The FIG. 3 circuit is an example of prior art that overcomes the staircase saturation problem associated with the FIG. 2 circuit. The two transformers are coupled inductors and energy storage devices that accommodate large DC currents, so that staircase saturation is not a problem. The FIG. 3 circuit can accomplish zero voltage switching under the right circumstances. The transitions are driven by the stored energy in the parallel inductor. The amount of energy stored in the parallel inductor must be large enough to charge/discharge the parasitic capacitors associated with the switches that are in transition. The current in the parallel inductor would have to be increased sufficiently to both provide current to drive the transition and current to drive the primary windings of the transformers. This is a particular problem at high line voltage where the energies required to drive the switching transition are greatest. For illustration, referring to FIG. 3, consider the condition in which switches S 1  and S 4  are on and switches S 2  and S 3  are off. Increasing current will flow from left to right through both the parallel inductor, L 1 , and the primary windings of the transformers. During this time current flows in the secondary winding of T 1  through D 1  and to the output capacitor and load resistor. Stored energy builds up in the core of transformer T 2 , but no current flows in the secondary winding of T 2 , since its secondary winding voltage reverse biases D 2 . After a time, switch S 1  is turned off and the stored energies in L 1  and T 2  drive the transition which can easily be made to be zero voltage. During the state which follows the connection point between T 1  and T 2  drops below ground potential and the T 1  primary voltage becomes equal to the sum of the voltages across T 2  and the switches, S 2  and S 4 . During the time that S 2  and S 4  are on, the current in L 1  remains relatively unchanged, dropping slightly, but the current in the primary windings drops towards zero as the current in the secondary windings shifts from T 1  to T 2 . The critical switching transition occurs when switch S 4  turns off. During the switching transition that follows L 1  must provide all of the energy to drive the transition, charging the output capacitance of S 4 , discharging the output capacitance of S 3  and providing charge to the other parasitic capacitances in the windings of each of the magnetic circuit elements and the D 1  diode, which becomes reverse biased during the transition. As secondary current shifts from D 1  to D 2  the current in the primary circuit reverses sign and flows from right to left. When the transition is complete the current in the primary winding will equal the magnetizing current in the primary winding of the T 1  transformer. In this discussion, and all the discussions that follow, the magnetizing current will mean the current in a coupled inductor winding that is substantially proportional to the field of magnetic induction that exists in the core of the coupled inductor. The magnetizing current in a coupled inductor may be referred to any winding of that coupled inductor in such a manner that the total stored magnetic energy in the core of the coupled inductor is equal to one half times the inductance of the winding, to which the magnetizing current is referred, times the square of the magnetizing current. With this definition of magnetizing current the magnetizing current will have both AC and DC components, in general. As the switching transition progresses the current required by the primary circuit from L 1  increases as the current provided by L 1  decreases. The rate of increase of current in the primary windings from right to left during the S 4  turn off transition depends on the line voltage and the resistance in the active section of the circuit. As the current in the primary windings increases from right to left much of the current and energy provided by the L 1  inductor is diverted to driving the load. In order for the current in the inductor L 1  to drive the load during the transition and also drive the transition the current provided by the inductor L 1  must be larger than the peak primary current and the energy stored must be sufficient that the current provided by L 1  is relatively invariant for the duration of the transition. As a result of the large current in L 1  the conduction losses in the four switches are substantially increased by the presence of L 1  and because of the substantial stored energy requirement the inductor L 1  adds additional cost, weight and volume to the converter. 
     OBJECTS AND ADVANTAGES 
     An object of the invention is to accomplish zero voltage switching, with the addition of a single small magnetic circuit element specifically for driving switching transitions, and thereby to reduce semiconductor switching power losses. 
     Another object is to provide an isolated converter which is relatively simple and is capable of delivering high output power. 
     Another object is to eliminate the possibility of transformer staircase saturation. 
     Another object is to provide a converter design with minimal snubber requirements and superior EMI performance. 
     Another object is to provide a simple resonant transition converter design that can be readily used with the single frequency, pulse width modulated or phase shift modulated, controller integrated circuits. 
     Another object is to provide a high power conversion scheme with reduced conduction losses. 
     Another object is to provide a resonant switching transition mechanism which can be designed to provide zero voltage switching over the full range of line voltage and load conditions. 
     Another object of this invention is to provide a high efficiency zero voltage switching power converter design that can be extended to multiple isolated outputs. 
     Another objective is to provide a high power bridge converter that neither needs nor benefits from the addition of a capacitor placed in series with a primary winding of a coupled inductive magnetic element. 
     Another objective is to provide a converter with superior output ripple performance. 
     Another object is to provide a resonant switching transition mechanism with two magnetic circuit elements, which store the energy transferred to the load during a switching cycle and with a small series inductance which enables the primary circuit current direction to be maintained through the switching transitions thereby enabling the resonant transitions. 
     Further objects and advantages of my invention will become apparent from a consideration of the drawings and ensuing description. 
     These and other objects of the invention are provided by a novel circuit technique that uses two coupled inductors as both energy storage devices and isolation mechanisms, a small inductance placed in series with the coupled inductors, and a secondary side switch for each coupled inductor and each output. The zero voltage switching transitions are accomplished by using the small series inductance to maintain the primary current direction through the switching transitions. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The present invention is illustrated by reference to the drawings in which like reference numerals refer to like elements of the invention. 
     FIG. 1 illustrates a circuit schematic drawing of a frequency modulated, zero voltage switching power converter. 
     FIG. 2 illustrates a circuit schematic drawing of a phase shift modulated, full bridge, zero voltage switching, forward converter. 
     FIG. 3 illustrates a circuit schematic drawing of a phase shift modulated, full bridge, zero voltage switching, dual opposed interleaved coupled inductor buck converter. 
     FIG. 4 illustrates a circuit schematic drawing of a soft switching, phase shift modulated, full bridge, dual opposed interleaved coupled inductor buck converter according to the subject invention. 
     FIG. 5 a  illustrates the wave form of the voltage applied to the primary circuit. 
     FIG. 5 b  illustrates the wave form of the magnetizing current in the first coupled inductor. 
     FIG. 5 c  illustrates the wave form of the magnetizing current in the second coupled inductor. 
     FIG. 5 d  illustrates the wave form of the current in the primary circuit. 
     FIG. 5 e  illustrates the wave forms of the currents in the secondary windings of the coupled inductors. 
     FIG. 6 illustrates an initial condition and second off state of the FIG. 4 converter. 
     FIG. 7 illustrates a first phase of a first turn on switching transition of the FIG. 4 converter. 
     FIG. 8 illustrates a second phase of a first turn on switching transition of the FIG. 4 converter. 
     FIG. 9 illustrates a third phase of a first turn on switching transition of the FIG. 4 converter. 
     FIG. 10 illustrates a fourth phase of a first turn on switching transition of the FIG. 4 converter. 
     FIG. 11 illustrates a first on state of the FIG. 4 converter. 
     FIG. 12 illustrates a first phase of a first turn off transition of the FIG. 4 converter. 
     FIG. 13 illustrates a second phase of a first turn off transition of the FIG. 4 converter. 
     FIG. 14 illustrates a first off state of the FIG. 4 converter. 
     FIG. 15 illustrates a first phase of a second turn on transition of the FIG. 4 converter. 
     FIG. 16 illustrates a second phase of a second turn on transition of the FIG. 4 converter. 
     FIG. 17 illustrates a third phase of a second turn on transition of the FIG. 4 converter. 
     FIG. 18 illustrates a fourth phase of a second turn on transition of the FIG. 4 converter. 
     FIG. 19 illustrates a second on state of the FIG. 4 converter. 
     FIG. 20 illustrates a first phase of a second turn off transition of the FIG. 4 circuit. 
     FIG. 21 illustrates a second phase of a second turn off transition of the FIG. 4 circuit. 
     FIG. 22 illustrates the power converter of FIG. 4 with diodes used as secondary switches and power mosfets used as primary switches. 
     FIG. 23 illustrates a circuit schematic drawing of a soft switching, active reset, pulse width modulated, dual opposed interleaved coupled inductor buck converter according to the subject invention. 
     FIG. 24 a  illustrates the wave form of the voltage applied to the primary circuit of the FIG. 23 converter. 
     FIG. 24 b  illustrates the wave form of the T 1  coupled inductor magnetizing current of the FIG. 23 converter. 
     FIG. 24 c  illustrates the wave form of the T 2  coupled inductor magnetizing current of the FIG. 23 converter. 
     FIG. 24 d  illustrates the wave form of the primary circuit current of the FIG. 23 converter. 
     FIG. 24 e  illustrates the wave forms of the secondary winding currents of the FIG. 23 converter. 
     FIG. 25 a  illustrates the wave form of the S 1  primary switch current of the FIG. 23 converter. 
     FIG. 25 b  illustrates the wave form of the S 2  primary switch current of the FIG. 23 converter. 
     FIG. 25 c  illustrates the wave form of the S 3  secondary switch current of the FIG. 23 converter. 
     FIG. 25 d  illustrates the wave form of the S 4  secondary switch current of the FIG. 23 converter. 
     FIG. 26 illustrates an on state of the FIG. 23 converter. 
     FIG. 27 illustrates a first phase of a turn off transition of the FIG. 23 converter. 
     FIG. 28 illustrates a second phase of a turn off transition of the FIG. 23 converter. 
     FIG. 29 illustrates a third phase of a turn off transition of the FIG. 23 converter. 
     FIG. 30 illustrates a fourth phase of a turn off transition of the FIG. 23 converter. 
     FIG. 31 illustrates an off state of the FIG. 23 converter. 
     FIG. 32 also illustrates the off state of the FIG. 23 converter. 
     FIG. 33 illustrates a first phase of a turn on transition of the FIG. 23 converter. 
     FIG. 34 illustrates a second phase of a turn on transition of the FIG. 23 converter. 
     FIG. 35 illustrates a third phase of a turn on transition of the FIG. 23 converter. 
     FIG. 36 illustrates a fourth phase of a turn on transition of the FIG. 23 converter. 
     FIG. 37 illustrates a fifth phase of a turn on transition of the FIG. 23 converter. 
     FIG. 38 illustrates an embodiment of the FIG. 23 converter in which power mosfets are used as primary switches and diodes are used as secondary switches. 
     FIG. 39 illustrates an embodiment of the FIG. 23 converter in which rectifier diodes are added to clamp ringing associated with the resonance of the L 1  inductor and the secondary diode parasitic capacitance. 
     FIG. 40 illustrates an embodiment of the FIG. 4 converter in which rectifier diodes are added to clamp ringing associated with the resonance of the L 1  inductor and the secondary diode parasitic capacitance. 
     FIG. 41 illustrates an embodiment of the FIG. 4 circuit in which the small inductance is placed in series with the secondary windings of the coupled inductors. 
     FIG. 42 illustrates an embodiment of the FIG. 23 circuit in which the small inductance is placed in series with the secondary winding of one of the coupled inductors. 
     FIG. 43 illustrates a magnetic circuit element construction in which both coupled inductors are combined on a single magnetic core. 
     FIG. 44 illustrates a magnetic circuit element construction in which both coupled inductors are combined on a single magnetic core and the primary winding of the two coupled inductors are combined in a single primary winding. 
     FIG. 45 illustrates an embodiment of the FIG. 4 circuit which uses the magnetic circuit element structure of FIG.  43 . 
     FIG. 46 illustrates an embodiment of the FIG. 4 circuit which uses the magnetic circuit element structure of FIG.  44 . 
     FIG. 47 illustrates an embodiment of the FIG. 23 circuit which uses the magnetic circuit element structure of FIG.  43 . 
     FIG. 48 illustrates an embodiment of the FIG. 23 circuit which uses the magnetic circuit element structure of FIG.  44 . 
     FIG. 49 illustrates an embodiment of the FIG. 4 circuit with an LC circuit which increases the available energy for driving the resonant transitions when used with phase shift modulation. 
     FIG. 50 illustrates an embodiment of the FIG. 4 circuit with an LC circuit which increases the available energy for driving the resonant transitions when used with pulse width modulation. 
     FIG. 51 illustrates an embodiment of the FIG. 23 circuit with an LC circuit which increases the available energy for driving the resonant transitions. 
     
       
         
               
             
               
               
               
               
               
             
           
               
                   
               
               
                 Reference Numerals 
               
               
                   
               
             
             
               
                   
               
             
          
           
               
                   
                 100 
                 DC input power source 
                 101 
                 positive terminal 
               
               
                   
                 102 
                 negative terminal 
                 103 
                 lead 
               
               
                   
                 104 
                 lead 
                 105 
                 node 
               
               
                   
                 106 
                 node 
                 107 
                 input capacitor 
               
               
                   
                 109 
                 lead 
                 110 
                 lead 
               
               
                   
                 111 
                 node 
                 112 
                 node 
               
               
                   
                 113 
                 capacitor 
                 114 
                 switch 
               
               
                   
                 115 
                 diode 
                 116 
                 capacitor 
               
               
                   
                 117 
                 switch 
                 118 
                 diode 
               
               
                   
                 119 
                 node 
                 120 
                 node 
               
               
                   
                 121 
                 lead 
                 122 
                 lead 
               
               
                   
                 123 
                 node 
                 128 
                 inductor 
               
               
                   
                 130 
                 coupled inductor 
                 131 
                 coupled inductor 
               
               
                   
                 134 
                 node 
                 135 
                 lead 
               
               
                   
                 136 
                 lead 
                 137 
                 node 
               
               
                   
                 138 
                 node 
                 139 
                 capacitor 
               
               
                   
                 140 
                 switch 
                 141 
                 diode 
               
               
                   
                 142 
                 capacitor 
                 143 
                 switch 
               
               
                   
                 144 
                 diode 
                 145 
                 node 
               
               
                   
                 146 
                 node 
                 147 
                 lead 
               
               
                   
                 148 
                 lead 
                 155 
                 switch 
               
               
                   
                 156 
                 switch 
                 161 
                 node 
               
               
                   
                 162 
                 node 
                 163 
                 capacitor 
               
               
                   
                 164 
                 load 
                 200 
                 DC input power source 
               
               
                   
                 201 
                 positive terminal 
                 202 
                 negative terminal 
               
               
                   
                 203 
                 lead 
                 204 
                 lead 
               
               
                   
                 205 
                 node 
                 206 
                 node 
               
               
                   
                 207 
                 capacitor 
                 208 
                 lead 
               
               
                   
                 209 
                 capacitor 
                 211 
                 node 
               
               
                   
                 212 
                 node 
                 213 
                 capacitor 
               
               
                   
                 214 
                 switch 
                 215 
                 diode 
               
               
                   
                 216 
                 capacitor 
                 217 
                 switch 
               
               
                   
                 218 
                 diode 
                 219 
                 node 
               
               
                   
                 220 
                 node 
                 221 
                 lead 
               
               
                   
                 222 
                 lead 
                 223 
                 node 
               
               
                   
                 228 
                 inductor 
                 230 
                 coupled inductor 
               
               
                   
                 231 
                 coupled inductor 
                 253 
                 switch 
               
               
                   
                 254 
                 switch 
                 261 
                 node 
               
               
                   
                 262 
                 node 
                 263 
                 capacitor 
               
               
                   
                 264 
                 load 
               
               
                   
                   
               
             
          
         
       
     
    
    
     SUMMARY 
     The subject invention uses a primary switching network that provides alternating bidirectional voltage to a pair of coupled inductors, which are also magnetic energy storage elements. A small series inductance is placed in series with the two coupled inductors. There is at least one secondary side switch for each coupled inductor secondary winding, a primary side energy storage and filter capacitor, and a secondary side energy storage and filter capacitor placed in parallel with the load. The zero voltage switching is accomplished in this converter by the current in the primary winding which is maintained in direction through the transition by the small inductance in series with the coupled inductors. 
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Referring to FIG. 4, there is shown a series type power processing topology. The circuit employs a source of substantially DC voltage, a primary switching network consisting of four switches in a full bridge arrangement that generates alternating bidirectional voltage pulses, a pair of coupled inductors arranged so that the primary windings of the two coupled inductors are in series and their secondary windings are connected at a point that is also common to a load and output filter capacitor, a small inductor placed in series with the primary windings of the coupled inductors, a pair of power switches, one in series with each coupled inductor secondary winding and connected to each other at a point which is also common to the output filter capacitor and the load. For purposes of the operational state analysis, it is assumed that the filter capacitor is sufficiently large that the voltage developed across the capacitor is approximately constant over a switching interval. Also for purposes of the operational state analysis, it is assumed that the input DC voltage source has sufficiently low source impedance that the voltage developed across the input DC voltage source is approximately constant over a switching interval. It will be assumed that the parasitic capacitors that parallel the switches are small and their effects can be ignored, except during the switching transitions. It will be assumed that the coupled inductor windings are coupled with a coupling coefficient of unity. It will be assumed that diodes are ideal and have no leakage and no forward voltage drop. It will finally be assumed that the power switches are ideal; that is, lossless and able to carry current in either direction. Moreover it will be recognized by one skilled in the art that, while only a single output version is considered in this analysis, multiples of voltages may be obtained through the addition of windings, switches, diodes, and capacitors operated as herein to be described. 
     STRUCTURE 
     The circuit structure is illustrated in FIG. 4. A positive terminal  101  of a source of DC potential, V IN ,  100  is connected to a lead  103 . The lead  103  is connected to a node  105 . A negative terminal  102  of source  100  is connected to a lead  104  which is connected to a node  106 . A first terminal of an input capacitor  107  is connected to the node  105 . A second terminal of the input capacitor  107  is connected to the node  106 . The node  105  is connected to a lead  109 . The lead  109  is connected to a node  111 . The node  106  is connected to a lead  110  which is connected to a node  112 . Node  111  is connected to a lead  135 . Lead  135  is connected to a node  137 . Node  112  is connected to a lead  136 . Lead  136  is connected to a node  138 . A first terminal of a capacitor  113  is connected to node  111 . A second terminal of the capacitor  113  is connected to a node  119 . A first terminal of a switch  114  is connected to node  111 . A second terminal of the switch  114  is connected to the node  119 . A cathode terminal of a diode  115  is connected to the node  111 . An anode terminal of the diode  115  is connected to the node  119 . A first terminal of a capacitor  116  is connected to the node  112 . A second terminal of the capacitor  116  is connected to a node  120 . A first terminal of a switch  117  is connected to the node  112 . A second terminal of the switch  117  is connected to the node  120 . An anode terminal of a diode  118  is connected to the node  112 . A cathode terminal of the diode  118  is connected to the node  120 . The node  119  is connected to a lead  121 . The node  120  is connected to a lead  122 . Lead  121  and lead  122  are connected at a node  123 , labeled B in FIG. 4. A first terminal of a capacitor  139  is connected to the node  137 . A second terminal of the capacitor  139  is connected to a node  145 . A first terminal of a switch  140  is connected to the node  137 . A second terminal of the switch  140  is connected to the node  145 . A cathode terminal of the diode  141  is connected to the node  137 . An anode terminal of the diode  141  is connected to the node  145 . A first terminal of a capacitor  142  is connected to the node  138 . A second terminal of the capacitor  142  is connected to a node  146 . A first terminal of a switch  143  is connected to the node  138 . A second terminal of the switch  143  is connected to the node  146 . An anode terminal of a diode  144  is connected to the node  138 . A cathode terminal of the diode  144  is connected to the node  146 . The node  145  is connected to a lead  147 . The node  146  is connected to a lead  148 . The lead  147  and the lead  148  are connected at a node  134 , labeled A in FIG. 4. A first terminal of an inductor  128  is connected to the node  123 . A second terminal of the inductor  128  is connected to an undotted terminal of the primary winding of a coupled inductor  130 . A dotted terminal of the primary winding of the coupled inductor  130  is connected to a dotted terminal of a primary winding of a coupled inductor  131 . An undotted terminal of the primary winding of the coupled inductor  131  is connected to the node  134 . 
     In the secondary circuit a dotted terminal of a secondary winding of the coupled inductor  130  is connected to a first terminal of a switch  155 . An undotted terminal of the secondary winding of the coupled inductor  130  is connected to a node  162 . A second terminal of the switch  155  is connected to a node  161 . A dotted terminal of a secondary winding of the coupled inductor  131  is connected to a first terminal of a switch  156 . An undotted terminal of the secondary winding of the coupled inductor  131  is connected to the node  162 . A second terminal of the switch  156  is connected to the node  161 . The node  161  is connected to a first terminal of a capacitor  163 . A second terminal of the capacitor  163  is connected to the node  162 . A first terminal of a load  164  is connected to the node  161 . A second terminal of the load  164  is connected to the node  162 . 
     OPERATION 
     It is assumed in this analysis that the system has reached a settled operating condition. Except for the short, but finite, switching intervals there are four states of the circuit of FIG. 4, two on states and two off states. It is also assumed, for purpose of analysis, that the switching intervals between the states are approximately zero seconds and that capacitors  113 ,  116 ,  139 , and  142  are small and do not contribute significantly to the operation of the converter, except during the brief switching transitions. 
     In operation consider an initial condition, illustrated in FIG. 6, in which the voltage difference between points A and B, labeled V_AB in FIG. 5 a,  is substantially zero and current is flowing through the primary windings from B towards A. The current flowing in the primary windings of coupled inductors  130  and  131 , labeled I_AB in FIG. 5 d,  is negative. During this initial condition, the switches  114  and  140  are closed (on) and switches  117  and  143  are open (off) and switch  156  is closed (on) and switch  155  is open (off). During this initial condition, coupled inductors  130  and  131  each have a substantial amount of stored energy, but the magnetizing currents of coupled inductors  130  and  131  referred to their primary windings are oppositely directed. Let us define that the magnetizing current is flowing in a positive direction when the current flows from the undotted terminal, through the winding, to the dotted terminal. For coupled inductor  130  the secondary current is zero and the primary current is equal to its magnetizing current, and the magnetizing current of coupled inductor  130 , referred to the primary winding, labeled I_MT 1  in FIG. 5 b,  is positive, since the current is flowing through the coupled inductor  130  primary winding from the undotted terminal to the dotted terminal. During the initial condition, the current I_AB is equal in magnitude and direction to the magnetizing current, referred to the primary winding, of coupled inductor  130 . During the initial condition, the magnetizing current, referred to the primary winding, of coupled inductor  131 , is in the positive direction and directed in opposition to the current I_AB. The total current in the primary winding of coupled inductor  131  is equal to the current I_AB and is composed of two components. One component of the total primary winding current of coupled inductor  131  is its magnetizing current and the other component is the induced current from the secondary winding of coupled inductor  131 . The induced current component of the primary winding current of coupled inductor  131  is equal to the current flowing in the secondary winding of coupled inductor  131  multiplied by the ratio of the secondary winding turns of coupled inductor  131  to the primary winding turns of coupled inductor  131 . Alternately, the current flowing in the secondary winding of coupled inductor  131  is equal to the vector sum of (1) the total current flowing in the primary winding of coupled inductor  131  multiplied by its turns ratio and (2) the magnetizing current of coupled inductor  131 , referred to the secondary winding. In this case the vector sum is equal to the sum of the magnitudes of the total primary winding current and the magnetizing current of coupled inductor  131  so that, if we assume a turns ratio of unity, the secondary winding current of coupled inductor  131  will be larger in magnitude than the primary winding current of inductor  131 . Another way to understand this situation is to consider that coupled inductor  131  magnetizing current is flowing in its secondary winding and that the magnetizing current component of the secondary winding current is supplemented by a contribution of current, induced from the primary winding, equal to the total primary winding current multiplied by the ratio of primary winding turns to secondary winding turns. During the initial condition, the dotted terminal of each coupled inductor winding is positive with respect to the negative terminal of each winding. The voltage V_AB is zero during the initial condition. Therefore, the primary winding voltage of coupled inductor  130  must be equal in magnitude to the primary winding voltage of coupled inductor  131  minus the winding voltage of the inductor L 1 . The magnitude of the voltage on the secondary winding of coupled inductor  131  is equal to the output voltage of the circuit. The magnitude of the voltage of the primary winding of coupled inductor  131  is equal to the secondary winding voltage of coupled inductor  131  multiplied by the ratio of the primary winding turns of coupled inductor  131  to the secondary winding turns of coupled inductor  131 . The magnitude of the voltage of the primary winding of coupled inductor  130  is equal to the magnitude of the primary winding voltage of coupled inductor  131  minus the voltage of the inductor  128 . The magnitude of the secondary winding voltage of coupled inductor  130  is equal to the magnitude of the primary winding voltage of coupled inductor  130  multiplied by the ratio of the primary winding turns of coupled inductor  130  to the secondary winding turns of coupled inductor  130 . During the initial condition, the winding voltages of inductors  128 ,  130 , and  131  are directed so that their stored energies and their magnetizing currents are decreasing. The circuit currents are illustrated in FIGS. 5 b  through  5   e.  At a time determined by the control circuit, the primary switching network changes state, so that switch  114  is opened (turned off). This condition is illustrated in FIG.  7 . The current in the primary windings, flowing from B to A, is maintained by the stored energy in the inductor  128  and this current charges capacitor  113  and discharges capacitor  116 . Shortly after switch  114  is opened switch  155  is closed. While the charging and discharging of capacitors  113  and  116 , respectively, is taking place, the voltage at B, which is node  123 , is falling and, eventually, diode  118  becomes forward biased and begins to conduct, as illustrated in FIG.  8 . As diode  118  is conducting, the voltage at B is clamped to a voltage substantially equal to the voltage at the negative terminal of V_IN. Shortly after diode  118  begins to conduct, switch  117  is closed (turned on) at substantially zero voltage, as illustrated in FIG.  9 . After switch  117  is closed, the current in the primary windings changes rapidly due to the large voltage applied to inductor  128  and reverses direction, as illustrated in FIG.  10 . At the same time the current in the secondary winding of coupled inductor  131  falls rapidly as the current in the secondary winding of coupled inductor  130  increases rapidly. When the current in the secondary winding of coupled inductor  131  drops to zero switch  156  is turned off (opened), as illustrated in FIG.  11 . When switch  156  is turned off the primary winding voltage of coupled inductor  131  and the inductor  128  voltage change rapidly. FIG. 11 illustrates a first on state in which the magnetizing current and stored energy in the coupled inductor  131  increase provided by energy from source V_IN  100 . During the first on state, current flows from right to left in FIG.  4  and from A to B in inductor  128  and the primary windings of coupled inductors  130  and  131 . During the first on state, the voltage V_AB is positive and the current I_AB is positive, as illustrated in FIG. 5 a  and FIG. 5 d,  respectively. The current in the secondary winding of coupled inductor  130  has two components. One component of the current in the secondary winding of coupled inductor  130  is due to the decreasing magnetizing current in coupled inductor  130  and the second component is due to the increasing reflected (induced) primary winding current. The magnetizing current in the secondary winding of coupled inductor  130  is decreasing because the dotted terminals of the windings of coupled inductor  130  are positive with respect to the undotted terminals of the windings of coupled inductor  130 . The reflected (induced) primary current is increasing because the primary current of coupled inductor  130  is equal to the primary magnetizing current of coupled inductor  131 , which is increasing. The magnetizing current of coupled inductor  131  is increasing because the undotted terminals of the windings of coupled inductor  131  are positive with respect to the dotted terminals of the windings of coupled inductor  131 . The result of both increasing and decreasing components of secondary winding current of coupled inductor  131  is a secondary winding current of coupled inductor  131  that increases. During the first on state, the secondary winding voltage of coupled inductor  130  is equal to the output voltage. The voltage of the primary winding of coupled inductor  130  is equal to the output voltage multiplied by the ratio of the primary turns to the secondary turns of coupled inductor  130 . The voltage applied to the primary winding of coupled inductor  131  is equal to the source voltage, V_IN, minus the voltage of the primary winding of coupled inductor  130  minus the voltage applied to inductor  128 . During the first on state both the stored energy and the magnetizing current in coupled inductor  131  increase. Since the inductor  128  and coupled inductor  131  are in series the current in inductor  128  is equal to the current in coupled inductor  131  and the energy stored in inductor  128  is increasing during the first on state. The first on state comes to an end at a time determined by the control circuit when switch  140  is opened (turned off), as illustrated in FIG.  12 . The stored energies in inductor  128  and coupled inductor  131  maintains the current in the primary winding of coupled inductors  130  and  131  as capacitor  139  is charged and capacitor  142  is discharged. As capacitor  142  discharges, the voltage at A node  134  drops toward the voltage of the negative terminal of V_IN  102 . When the voltage at A  134  reaches the voltage of the negative terminal  102  of V_IN, diode  144  begins to conduct, as illustrated in FIG. 13, and clamps the voltage at A node  134 , preventing the voltage at A from dropping further. After diode  144  begins to conduct, switch  143  is closed (turned on), as illustrated in FIG.  14 . The system, as illustrated in FIG. 14, is now in a first off state, in which the primary winding current is flowing from right to left or from A to B. During the first off state the magnetizing currents and stored energies in inductors  128 ,  130 , and  131  decrease. During the first off state the secondary winding current of coupled inductor  130  decreases. The voltage across the secondary winding of coupled inductor  130  is equal to the output voltage. The voltage across the primary winding of coupled inductor  130  is equal to the secondary winding voltage of coupled inductor  130  multiplied by the ratio of the primary turns to the secondary turns of coupled inductor  130 . At a time determined by the control circuit, switch  117  is turned off, as illustrated in FIG.  15 . Immediately thereafter switch  156  is turned on (closed). The stored energy in inductor  128  maintains the primary winding currents of coupled inductors  130  and  131  as capacitor  116  is charged and capacitor  113  is discharged. The voltage at B, node  123 , rises as capacitors  116  and  113  charge and discharge, respectively, until the voltage at B reaches the voltage at the positive terminal  101  of V_IN, at which time diode  115  begins to conduct, as illustrated in FIG. 16, and clamps the voltage at B, preventing the voltage at B from continuing to rise above the voltage at the positive terminal of V_IN. Soon after diode  115  begins to conduct, switch  114  is closed (turned on), as illustrated in FIG.  17 . After switch  114  is closed, the current in switch  156  increases rapidly, as the current in switch  155  decreases rapidly. During this time the primary current magnitude will drop rapidly and change sign, as illustrated in FIG.  19 . When the switch  155  drops to zero the switch  155  is opened (turned off), as illustrated in FIG.  19 . The opening of switch  155  marks the beginning of a second on state. During the second on state, illustrated in FIG. 19, the secondary winding voltage of coupled inductor  131  is equal to the output voltage. The primary winding voltage of coupled inductor  131  is equal to the output voltage multiplied by the ratio of the primary turns to the secondary turns of coupled inductor  131 . The primary winding voltage of coupled inductor  130  is equal to the input DC voltage, V_IN, minus the primary winding voltage of coupled inductor  131  minus the inductor  128  voltage. During the second on state, the magnetizing current and stored energy of coupled inductor  130  increase. The secondary winding current of coupled inductor  131  has two components. One component of the secondary winding current of coupled inductor  131  is equal to the decreasing magnetizing current, referred to the secondary winding, of coupled inductor  131 . The second component of the secondary winding current of coupled inductor  131  is equal to the increasing reflected (induced) primary current in coupled inductor  130 . The primary current of coupled inductor  131  is equal to the increasing magnetizing current of coupled inductor  130 . The net current in the secondary winding of coupled inductor  131  will be increasing, during the second on state. During the second on state, the voltage V_AB is negative and the current I_AB is also negative since the voltage at B is positive with respect to the voltage at A and the current in the primary windings of coupled inductors  130  and  131  is flowing from B towards A. During the second on state, stored energy in coupled inductor  130  increases as the magnetizing current in coupled inductor  130  increases. At a time determined by the control circuit, switch  143  is opened (turned off), as illustrated in FIG.  20 . The stored energy in coupled inductor  130  maintains the current in the primary windings of coupled inductors  130  and  131  and this current charges capacitor  142  and discharges capacitor  139 . As capacitors  142  and  139  charge and discharge, respectively, the voltage at A rises, until diode  141  becomes forward biased, as illustrated in FIG. 21, and clamps the voltage at A to the voltage at the positive terminal of V_IN. Soon after diode  141  begins to conduct, switch  140  is closed (turned on), as illustrated in FIG.  6 . The system is now in a second off state and the conditions are the same as the initial conditions, as illustrated in FIG. 6, and the process described above repeats continuously. The drive signal to switch  117  is inverted, with respect to the drive signal of switch  114 , and the drive signal to switch  143  is inverted, with respect to the drive signal to switch  140 , except that there are short time intervals, during which the resonant transitions take place, in which the switches are off simultaneously. Break-before-make switching, with a short, but finite, switching interval, which may be of the order of approximately 20 to 2000 nanoseconds, is adopted in the implementation of zero voltage switching. Each of the switching transitions of switches  114 ,  117 ,  140 , and  143  are zero voltage so that to a first order approximation the switching losses of the four primary switches are eliminated. 
     RELATED EMBODIMENTS 
     FIG. 22 illustrates an embodiment of the subject invention, in which the secondary switches are implemented using diodes and the primary switches are implemented using power mosfets. The power mosfets contain the switch, as a semiconductor field effect transistor switch, the diode as the intrinsic body drain diode of the power mosfet, and the capacitor as the mosfet output capacitance. The diode provides a natural turn off at the time that the current drops to zero, as needed. 
     FIG. 40 shows an embodiment related to the FIG. 22 embodiment in which a pair of diodes D 3  and D 4  are added to clamp potential ringing due to the interaction of L 1  with the parasitic junction capacitances of D 1  and D 2 . 
     The control block suggests that either phase shift modulation or pulse width modulation may be used. There is nothing that would prevent either control method from being employed in this invention. The phase shift modulation approach has been described here as an example of one control approach but pulse width modulation of the primary switches, using the described circuit structure and operating mechanism, must be considered as included within the claims of this invention. The invention does nothing to preclude the use of pulse width modulation, as an alternative method of control. It is well known to anyone skilled in the art of power conversion that any full bridge switching structure can be modulated by either phase shift modulation or pulse width modulation to obtain the desired control. With pulse width modulation both off states would use the same pair of switches and there would be three distinct switch states rather than four, as described with the phase shift modulation approach. 
     Another embodiment of the FIG. 4 structure is illustrated in FIG.  41 . In the FIG. 41 embodiment the series inductance is placed in the secondary circuit rather than the primary circuit. In the FIG. 41 embodiment two small inductors are required, one for each secondary winding. In the FIG. 41 embodiment the energy to drive the transitions is provided by the magnetizing energy of the coupled inductors and the effect of the series inductance is to maintain the primary current largely unchanged throughout the duration of the switching transition. The FIG. 41 embodiment is provided for completeness, but offers no advantage over the other embodiments already discussed. In fact, there are disadvantages to this approach: two inductors in the secondary circuit are required rather than one placed in the primary circuit, and there is no simple mechanism for clamping the ringing associated with the small series inductances and the parasitic capacitance of the output rectifiers as there is in the FIG. 40 embodiment. 
     FIG. 43 illustrates a magnetic circuit element structure in which two coupled inductors are wound on a single E core. Each outer leg is gapped. The center post of the E core provides a common flux return path (magnetic short circuit) for each coupled inductor, but there is no operational difference between the FIG. 43 structure and two separate and independent coupled inductor structures. FIG. 44 illustrates a magnetic circuit element structure in which both primary windings are combined into a single primary winding placed on the center post of an E core and the secondary windings are placed on the outer legs of the structure. The operation of the FIG. 44 structure is similar to the FIG. 43 structure. Depending on the direction and magnitude of voltage applied to the primary winding one or the other or both of the secondary windings are coupled to the primary winding. Here again the center post serves as the common flux return path for both outer legs which are gapped. The difference between the operation of the FIG. 44 structure and the FIG. 43 structure is that the FIG. 44 structure will, in general, have greater leakage inductance and lower coupling to the secondary windings, which will, in general, be less efficient and noisier than the FIG. 43 structure. 
     Another embodiment of the FIG. 4 circuit is shown in FIG.  45 . This embodiment employs the magnetic structure of FIG.  43 . 
     Another embodiment of the FIG. 4 circuit is shown in FIG.  46 . This embodiment employs the magnetic structure of FIG.  44 . 
     Another embodiment of the FIG. 4 circuit is shown in FIG.  49 . This embodiment adds a LC tank circuit that provides additional energy and current for driving the switching transitions. The LC tank circuit is helpful for applications with a wide line voltage range since the energy and current provided by the tank circuit increases as it is needed with increasing line voltage. The FIG. 49 embodiment is applicable to phase shift modulation where the critical switching transitions are confined to only one half bridge switching leg. 
     Another embodiment which is a variation of the FIG. 49 embodiment applicable to pulse width modulation control is shown in FIG.  50 . 
     Additional embodiments are realized by adding converter outputs. Additional converter outputs can be added by adding secondary windings to each coupled inductor and secondary switches for each secondary winding and output capacitors and loads for each additional output. Additional embodiments are realized by paralleling multiple converters of the type described herein with equally spaced phase. 
     STRUCTURE 
     The circuit structure is illustrated in FIG.  23 . The circuit structure is illustrated in FIG. 23. A positive terminal  201  of a source of DC potential, V IN ,  200  is connected to a lead  203 . The lead  203  is connected to a node  205 , labeled A in FIG. 23. A negative terminal  202  of source  200  is connected to a lead  204  which is connected to a node  206 . A first terminal of an input capacitor  207  is connected to the node  205 . A second terminal of the input capacitor  207  is connected to the node  206 . The node  205  is connected to a first terminal of a capacitor  209 . A second terminal of the capacitor  209  is connected to a node  211 . The node  206  is connected to a lead  208  which is connected to a node  212 . Node  211  is connected to a first terminal of a capacitor  213 . A second terminal of the capacitor  213  is connected to a node  219 . A first terminal of a switch  214  is connected to node  211 . A second terminal of the switch  214  is connected to the node  219 . A cathode terminal of a diode  215  is connected to the node  211 . An anode terminal of the diode  215  is connected to the node  219 . A first terminal of a capacitor  216  is connected to the node  212 . A second terminal of the capacitor  216  is connected to a node  220 . A first terminal of a switch  217  is connected to the node  212 . A second terminal of the switch  217  is connected to the node  220 . An anode terminal of a diode  218  is connected to the node  212 . A cathode terminal of the diode  218  is connected to the node  220 . The node  219  is connected to a lead  221 . The node  220  is connected to a lead  222 . Lead  221  and lead  222  are connected at a node  223 , labeled B in FIG. 23. A first terminal of an inductor  228  is connected to the node  205 . A second terminal of the inductor  228  is connected to an undotted terminal of the primary winding of a coupled inductor  230 . A dotted terminal of the primary winding of the coupled inductor  230  is connected to a dotted terminal of a primary winding of a coupled inductor  231 . An undotted terminal of the primary winding of the coupled inductor  231  is connected to the node  223 . 
     In the secondary circuit a dotted terminal of a secondary winding of the coupled inductor  230  is connected to a first terminal of a switch  253 . An undotted terminal of the secondary winding of the coupled inductor  230  is connected to a node  262 . A second terminal of the switch  253  is connected to a node  261 . A dotted terminal of a secondary winding of the coupled inductor  231  is connected to a first terminal of a switch  254 . An undotted terminal of the secondary winding of the coupled inductor  231  is connected to the node  262 . A second terminal of the switch  254  is connected to the node  261 . The node  261  is connected to a first terminal of a capacitor  263 . A second terminal of the capacitor  263  is connected to the node  262 . 
     A first terminal of a load  264  is connected to the node  261 . A second terminal of the load  264  is connected to the node  262 . 
     OPERATION 
     It is assumed in this analysis that the system has reached a settled operating condition. Except for the short, but finite, switching intervals there are two states of the circuit of FIG. 23, an on state and an off state. It is also assumed, for purpose of analysis, that the switching intervals between the states are approximately zero seconds in duration and that capacitors  213  and  216  are small and do not contribute significantly to the operation of the converter, except during the switching transitions. 
     It is also assumed that the capacitor  209  is large so that its voltage is invariant over the duration of a switching cycle. 
     In operation consider an initial condition, illustrated in FIG. 26, in which the voltage difference between points A, node  205 , and B, node  223 , labeled V_AB in FIG. 24 a,  is equal to the voltage V_IN and current is flowing from left to right or from A to B. The current flowing in the primary windings of coupled inductors  230  and  231 , labeled I_AB in FIG. 24 d,  is positive. During this initial condition, switch  217  is closed (on) and switch  214  is open (off). During this initial condition, coupled inductor  230  has a substantial amount of stored energy. During this initial condition, coupled inductor  231  also has a substantial amount of stored energy but, in general, the stored energy in coupled inductor  231  will be less than the stored energy in coupled inductor  230 . Let us define that the magnetizing current is flowing in a positive direction when the current flows from the undotted terminal through the winding to the dotted terminal. For coupled inductor  230 , during the initial condition, the secondary current is zero and the primary current is equal to its magnetizing current and the magnetizing current, referred to the primary winding, labeled I_MT 1  in FIG. 24 b,  is positive, since the current is flowing through its primary winding from the undotted terminal of its primary winding to the dotted terminal of its primary winding. During the initial condition, the current I_AB is equal in magnitude and direction to the magnetizing current, referred to the primary winding, of coupled inductor  230 . During the initial condition, the magnetizing current, referred to the primary winding, of coupled inductor  231  is decreasing. The total current in the primary winding of coupled inductor  231  is equal to the current I_AB. There are two components of the secondary winding current of coupled inductor T 2 . One component of the secondary winding current of coupled inductor T 2  is its magnetizing current, referred to the secondary winding, which is decreasing, and the other component is the induced current from its primary winding, which is increasing. By proper selection of component values the two slopes can be made equal and opposite so that the ripple current slope is zero. This property is very desirable since it reduces the size of the output capacitor. The induced current component of the secondary winding current of coupled inductor  231  is equal to the current flowing in the primary winding of coupled inductor  231  multiplied by the ratio of the primary winding turns to the secondary winding turns of coupled inductor  231 . During the initial condition, the secondary winding voltage of coupled inductor  231  is equal to the output voltage and the primary winding voltage of coupled inductor  231  is equal to the secondary winding voltage of coupled inductor  231  multiplied by the ratio of the primary winding turns to the secondary winding turns of coupled inductor  231 . The primary winding voltage of coupled inductor  230  is equal to the voltage of source V_IN minus the primary winding voltage of coupled inductor  231  minus the L 1  inductor  228  voltage. During the initial condition, the energy in inductors  230  and  228  increases. At a time determined by the control circuit the primary switching network changes state so that switch  217  is opened, as illustrated in FIG.  27 . The stored energy in coupled inductor  230  maintains the primary winding current, which is flowing from A towards B, and forces charge into C 2  capacitor  213 , into C 1  capacitor  216 , and into C 4  capacitor  209 . The voltage at B rises as capacitors C 1   216  and C 4   209  charge and capacitor C 2   213  discharges. Since the capacitance of C 4   209  is much larger than the capacitance of C 1   216  and C 2   213  we can assume that the voltage applied to C 4   209  remains invariant throughout the switching transition. When the voltage at B, node  223 , reaches the voltage at the positive terminal of V_IN  201 , switch  253  is closed (turned on), as illustrated in FIG.  28 . The voltage at B, node  223 , continues to rise and the current in switch  254  falls slightly and the current in the primary windings of coupled inductors  230  and  231  falls slightly as the current in switch  253  rises slightly. While switches  253  and  254  are closed, the output current begins to transfer from switch  254  to switch  253 . The voltage at B, node  223 , continues to rise, driven by the stored energy in inductor L 1   228  and coupled inductor  231 , until the diode  215  is forward biased, as illustrated in FIG.  29 . Soon thereafter, switch  214  is turned on at zero voltage, as illustrated in FIG.  30 . Now the current is transferring rapidly from  254  to  253  and the primary current is dropping rapidly. Soon the current in switch  254  drops to zero and switch  254  is opened (turned off), as illustrated in FIG.  31 . When the  254  switch is opened the primary current is equal to the magnetizing current of coupled inductor  231 , referred to its primary winding. FIG. 31 shows the operating condition for an off state. At the beginning of the off state the primary current is flowing from A towards B and the primary current is equal in magnitude to the magnetizing current of coupled inductor  231 , which is negative and increasing, i.e., becoming more positive as time progresses, as illustrated in FIG. 24 c.  By examining FIG. 31 one can see that the voltage on capacitor  209  is equal to the sum of the primary winding voltages of inductors  228 ,  230 , and  231 . The voltage on the secondary winding of coupled inductor  230  is equal to the converter&#39;s output voltage. The voltage on the primary winding of coupled inductor  230  is equal to the output voltage times the ratio of the primary winding turns to the secondary winding turns of coupled inductor  230 . The primary winding voltage of coupled inductor  231  is equal to the voltage of capacitor  209  minus the primary winding voltage of coupled inductor  230  minus the inductor  228  voltage. The voltage applied to the primary winding of coupled inductor  231  is directed so that the magnetizing current in coupled inductor  231  becomes more positive with the passage of time and the magnetizing current in coupled inductor  231  passes through zero and becomes positive, as illustrated in FIGS. 24 c  and  32 . When the magnetizing current in coupled inductor  231  has increased to a value approximately equal to the negative of its value at the beginning of the off state, switch  214  is opened (turned off). When switch  214  is turned off the energy stored in inductors  231  and  228  maintains the primary winding current, which is now flowing from B towards A, charging capacitor  213  and discharging capacitors  209  and  216 , as illustrated in FIG.  33 . The voltage at B drops during this time, until the voltage at B reaches the voltage at the positive terminal of V_IN. At this point switch  254  is closed at zero voltage, as illustrated in FIG.  34 . The voltage at B continues to fall driven by the stored energy in inductor  228 , until diode  218  becomes forward biased, as illustrated in FIG.  35 . Shortly after diode  218  begins to conduct switch  217  is closed (turned on), as illustrated in FIG.  36 . During this time the primary current and the secondary currents are changing rapidly as the secondary current transfers from  253  to  254  and the primary current drops in magnitude. Shortly after switch  217  is closed the primary current changes sign as illustrated in FIG.  37 . When the current in  253  drops to zero switch  253  is turned off (opened), as illustrated in FIG. 26, which also illustrates the initial condition. The initial condition is the on state of the converter. At this point a full cycle of operation has been completed and the initial conditions are again established. During the full operating cycle both of the primary switches were turned on and turned off at substantially zero voltage, eliminating first order switching losses for the two primary switches. 
     RELATED EMBODIMENTS 
     FIG. 38 shows an implementation of the circuit of FIG. 23 in which the secondary switches are diodes and the primary switches are power mosfets which intrinsically contain a capacitor and a diode in the form of the mosfet output capacitance and the body drain diode, respectively. 
     FIG. 39 shows an implementation in which diodes D 3  and D 4  are added to clamp ringing associated with the L 1  inductor and the parasitic capacitances of D 1  and D 2 . 
     FIG. 42 shows an embodiment in which the small inductor is placed in series with the D 2  diode. In this embodiment no inductance is placed in series with the T 1  coupled inductor. During the transition from the on state to the off state the series inductor of the FIG. 23 circuit is not relied upon to accomplish the zero voltage transition since the magnetizing current of the T 2  coupled inductor is directed to drive the transition, so an inductor placed in series with the T 1  coupled is not required to accomplish a zero voltage transition for the turn off transition. For the turn on transition the magnetizing current of the T 1  coupled inductor is, in general, directed to oppose the turn on transition. In this case the inductor placed in series with the T 2  coupled inductor is required to maintain the primary current through the turn on transition in order to accomplish zero voltage switching for the turn on transition. The FIG. 39 embodiment is preferred over the FIG. 42 embodiment since the FIG. 39 embodiment provides a mechanism to clamp the ringing associated with the resonance associated with the small series inductor and the parasitic capacitances of the output diodes. 
     Another embodiment is shown in FIG.  47 . In the FIG. 47 embodiment the two coupled inductors are replaced by a single integrated magnetic structure in which both coupled inductors are wound on a single core as illustrated in FIG.  43 . 
     Another embodiment related to the FIG. 47 embodiment is illustrated in FIG.  48 . In the FIG. 48 embodiment the applicable magnetic structure is illustrated in FIG.  44 . 
     Another embodiment is illustrated in FIG.  51 . In the FIG. 51 embodiment a LC tank circuit is added to provide additional stored magnetic energy for driving the switching transitions. 
     Additional embodiments can be realized by adding secondary windings to each coupled inductor and associated switches, capacitors, and loads to extend the concept to multiple outputs. Another embodiment is realized by paralleling interleaved converters of the type shown in FIG. 23 which share input power source, input capacitor, reset capacitor, output capacitor, and load. 
     CONCLUSION, RAMIFICATIONS, AND SCOPE OF INVENTION 
     Thus the reader will see that the power converters of the invention provide a mechanism which significantly reduces switching losses, has low component parts counts, and does not require high core losses or high conduction losses to accomplish zero voltage switching, relying on the finite rate of change of current associated with a small magnetic circuit element placed in series with a coupled inductor. 
     While my above description contains many specificities, these should not be construed as limitations on the scope of the invention, but rather as exemplifications of preferred embodiments thereof. Many other variations are possible. For example, other variations include power converters with more than one output; multi-phase, interleaved, parallel power converters with two or more parallel converter sections; power converters arranged in a bridged configuration for amplifier and inverter applications; power converters similar to those shown in the drawing but which integrate individual magnetic circuit elements onto a single magnetic core; power converters similar to those shown but which have instead high AC ripple voltages on the input filter capacitors; power converters, similar to those shown in the drawing, but where the DC input source is instead a varying rectified AC signal. Accordingly, the scope of the invention should be determined not by the embodiments illustrated, but by the appended claims and their legal equivalents.

Technology Category: 4