Patent Document

This is a Continuation of U.S. patent application Ser. No. 14/483,295 filed Sep. 11, 2014. 
    
    
     BACKGROUND 
     Generating broadband quadrature-modulated signals presents a number of challenges in achieving wide bandwidth and spectral purity, in shaping the waveform, in eliminating spurious components and non-linearities from the delivered signal, and in calibrating the quadrature modulation. Receiving and demodulating the signals presents similar challenges, particularly in cases of frequency conversion relating to image rejection and local oscillator (LO) leakage at the intermediate frequency (IF). Another issue involves quadrature imbalance at the local oscillator, where imbalances occurring at different points in the transmit/receive path are typically inseparable and are therefore not readily correctable. Thus, it would be desirable to have apparatus and methods for generating and receiving quadrature-modulated signals having not only wide bandwidth and spectral purity, but also featuring ease of calibration and rejection of undesirable artifacts in the signal. This goal is met by embodiments of the present invention. 
     SUMMARY 
     An embodiment of the present invention provides extremely wide-band signal-generating apparatus featuring multiple signal synthesizers and multiple quadrature modulators having independently-selectable configurations for flexible interconnections. Apparatus according to this embodiment allows convenient combination and isolation of different sections to enable convenient characterization of spectral components and filters for optimizing performance and rejection of spurious signal artifacts. 
     Another embodiment of the present invention provides quadrature modulators having internal digital filters to compensate for the frequency-dependencies of low-pass anti-aliasing filters. 
     A further embodiment of the present invention provides digital pre-processing apparatus for conditioning an input waveform to signal generation apparatus as disclosed herein. 
     Other embodiments of the present invention provide methods for self-calibration of analog and digital components of signal generating and receiving apparatus as disclosed herein. 
     Embodiments of the present invention are particularly well-suited to being incorporated within integrated circuits. 
     Further advantages offered by embodiments of the present invention include the ability for configurations to be adapted on the fly, and to be adaptively optimized according to the specific environment and operational settings. Embodiments of the present invention can thus be optimized in the various degrees of freedom (e.g. per frequency) for performance, spur rejection, interference resiliency, signal-to-noise ratio, bit error rate, and so forth. 
     It is understood that the present invention is not limited to the particular area of Radar and that embodiments of the invention are also applicable to other areas of the microwave signal field; including but not limited to: communications; radio frequency (RF) imaging; multiple input-multiple output (MIMO) communications and phased arrays; sensor-based applications (e.g. material analysis/monitoring); and test equipment implementation, such as vector network analyzers (VNA). 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The subject matter disclosed may best be understood by reference to the following detailed description when read with the accompanying drawings in which: 
         FIG. 1A  illustrates a signal generator with pre-corrected digital inputs according to an embodiment of the present invention. 
         FIG. 1B  illustrates a sideband selector switch for the signal generator of  FIG. 1A , according to a related embodiment of the present invention. 
         FIG. 2  illustrates a signal generator according to an embodiment of the present invention. 
         FIG. 3  illustrates a multiple signal generator according to an embodiment of the present invention. 
         FIG. 4  illustrates a transceiver according to an embodiment of the present invention. 
         FIG. 5  illustrates a quadrature receiver according to an embodiment of the present invention. 
         FIG. 6  illustrates a multistatic radar apparatus according to an embodiment of the present invention. 
         FIG. 7  illustrates a 3-channel MIMO transceiver according to an embodiment of the present invention. 
         FIG. 8  illustrates a spectral component measurement arrangement at the output of the signal generation block according to an embodiment of the present invention. 
         FIG. 9  illustrates a receiver-assisted spectral component measurement arrangement according to an embodiment of the present invention. 
         FIG. 10A  illustrates a symmetrized receiver-assisted spectral component measurement arrangement for characterization of a first quadrature modulation block according to the present invention. 
         FIG. 10B  illustrates a symmetrized receiver-assisted spectral component measurement arrangement for characterization of a second quadrature modulation block according to the present invention. 
         FIG. 11  illustrates a multi-module referenced based scaling arrangement according to an embodiment of the present invention. 
         FIG. 12  is a flowchart of a method of calibrating a signal generator according to an embodiment of the present invention. 
     
    
    
     For simplicity and clarity of illustration, elements shown in the figures are not necessarily drawn to scale, and the dimensions of some elements may be exaggerated relative to other elements. In addition, reference numerals may be repeated among the figures to indicate corresponding or analogous elements. 
     DETAILED DESCRIPTION 
       FIG. 1A  illustrates a signal generator  100  with pre-corrected digital inputs  181  and  183  according to an embodiment of the present invention. 
     In a quadrature modulation block  101 , digital-to-analog converters (DAC)  103  and  107 , respectively, receive digital inputs  181  and  183  and send analog signals corresponding thereto to anti-aliasing low-pass filters (LPF)  105  and  109 , respectively. Digital input  181  is a pre-corrected in-phase input I C , whereas digital input  183  is a pre-corrected quadrature input Q C . Anti-aliasing low-pass filters  105  and  109  in turn output signals to multiplicative mixers (“mixers”)  111  and  113 , respectively. A 90° splitter  115  receives a synthesized frequency from a synthesizer  121  and outputs two signals which are 90° out of phase, with the signal to mixer  113  lagging 90° behind the signal to mixer  111 . The mixed outputs from mixer  111  and mixer  113  are input to a summing unit  117 . 
     The output from quadrature modulation block  101  is input to a switch  133 A, which can be selectably switched to pass the direct output of quadrature modulation block  101  or the output of quadrature modulation block  101  mixed by a mixer  131  with a synthesized frequency from a synthesizer  123 . 
     Various embodiments of the invention feature switches configured in a manner similar to that of switch  133 A. Certain embodiments of the invention provide that these switches be independently selectably switchable. Independent switchability according to these embodiments of the invention not only provides versatility in configuring apparatus, but also provides benefits in calibration of the apparatus, as detailed below. 
     Quadrature modulation typically suffers from spurious image-frequency signal and from local oscillator feed-through. These imperfections can be significantly reduced by signal pre-compensation in the digital domain. The setting of the pre-compensation or pre-correction coefficients requires a feedback mechanism allowing the measurement of the above spurious signals. 
     Therefore, an embodiment of the present invention provides for pre-correction as follows. A numerically-controlled oscillator (NCO)  141  receives a frequency signal  143  to set the frequency f of the oscillator, and an initial phase signal  143  to set the initial phase φ 0 . Numerically-controlled oscillator  141  outputs two signals, a sine wave  147  sin (f, φ 0 ) and a cosine wave  149  cos (f, φ 0 ), which are input to a complex multiplier  151 , whose other inputs are an in-phase data stream  153  I data  (k) and a quadrature data stream  155  Q data (k). The complex product outputs of complex multiplier  151  are a desired in-phase data wave  157  I and a desired quadrature data wave  159  Q. However, in order to compensate for effects such as amplitude imbalance of quadrature modulation to be performed by quadrature modulation block  101 , a pre-correction is needed, which is furnished by a matrix multiplier  161 , containing filters  163 ,  165 ,  167  and  169  for single sideband (SSB) rejection. In addition, matrix multiplier  161  also corrects for local oscillator leakage with direct current offsets I DC  and Q DC  into summing mixers  177  and  175 , respectively. 
     Furthermore, in accordance with an embodiment of the present invention, digital filters  163 ,  165 ,  167 , and  169  feeding into summing mixers  171  and  173 , respectively, are incorporated into matrix multiplier  161  to compensate for the frequency-dependencies of anti-aliasing low pass filters  105  and  109 . The result, as previously noted, are pre-corrected in-phase input  181  I C  and pre-corrected quadrature input  183  Q C . 
       FIG. 1B  illustrates a sideband selector configuration switch  133 B according to a related embodiment of the present invention. Sideband selector configuration switch  133 B selectively switches between the direct output of quadrature modulation block  101  and either the upper sideband of the output of quadrature modulation block  101  mixed via mixer  131  with the output of synthesizer  123 , or the lower sideband thereof, as passed by an upper sideband filter  135  or a lower sideband filter  137 , respectively. 
     In the above descriptions, transmission signal generation is a hybrid configurable one/two conversion process as illustrated in  FIG. 1A . The different states reached under the topology depend on the setting of switch  133 A and are as follows:
         Direct conversion based on a frequency synthesizer  121 , which is directly modulated by wide-band quadrature modulator block  101 ;   Double conversion operation based on mixing between the output of quadrature modulator block  101  with synthesizer  123 .       

     This architecture inherently features an extremely wide frequency coverage (DC to 10s GHz) while maintaining low spurious signal content. In certain cases the synthesizer frequency range is increased by digital dividers. In these cases, for noise minimization and stability, it may be of interest to have the synthesizers operate at different frequencies. Digitally divided signals, however, typically have high spurious harmonic content. Operation over a multi-octave frequency range normally requires complicated re-configurable filters and filter banks to suppress these spurious signals. By heterodyne down-conversion of the direct modulated signal, wide frequency coverage can be achieved with the spurious signals lying out-of-band. 
     As the frequency coverage requirement widens, so does the coverage requirement from the synthesizers and direct modulators. Employing both direct and double conversions may relax the above requirement. For example, a quadrature modulator covering the range 4-8 GHz may be mixed with an additional 8-12 GHz synthesizer in order to cover the DC-4 GHz range, and with a 12-16 GHz synthesizer in order to cover the 8-12 GHz frequency range. Higher frequencies may be covered by using up-conversion rather than down-conversion. 
     Another benefit provided by embodiments of the present invention is the capability of arbitrarily modulating a wide-band waveform (as wide as the baseband) at any frequency within the frequency coverage. This permits the use of modulation schemes such as chirp/pseudo-random bit sequence (PRBS) for pulse compression in radar applications, communication constellations, and so forth. 
     Further use of the arbitrary digital modulation provided by embodiments of the present invention allows a fine-frequency offset in the digital domain. This permits coarser frequency steps in the synthesizers, improving their phase noise performance for the same frequency resolution requirement. 
     Another benefit provided by embodiments of the present invention is the ability to reach a certain output frequency via several different configurations. In a non-limiting example, by stepping the synthesizer to a higher frequency and correspondingly stepping the baseband frequency to a lower frequency the output frequency is unchanged. This is instrumental in producing a coherent frequency coverage across all synthesizer frequencies, even though it does not retain a specific phase over frequency change. 
       FIG. 2  illustrates a signal generator according to another embodiment of the present invention, where a second quadrature modulation block  203  is utilized to directly modulate synthesizer  123  to create the local oscillator for the second conversion. This enables a tradeoff of quadrature modulation imbalance versus phase noise to attain arbitrary frequency in generating the local oscillator for the conversion node. 
       FIG. 3  illustrates a multiple signal generator according to an embodiment of the present invention. Frequency synthesizer  301  feeds quadrature modulation blocks  303  and  305 , and frequency synthesizer  351  feeds quadrature modulation blocks  353  and  355 . Selector switches  311 ,  331 ,  361 , and  381  operate as previously described for selector switch  133 A ( FIG. 1A ), and selectably switch between the direct output of quadrature modulation blocks  303 ,  305 ,  353 , and  355  respectively, and outputs of mixers  313 ,  333 ,  363 , and  383 , respectively, all of which receive input from frequency synthesizer  391 . 
     As previously noted, various embodiments of the present invention provide selector switches  311 ,  331 ,  361 , and  381  to be independently switchable. 
     The arrangement illustrated in  FIG. 3  is useful in Radar communication systems where there is a need for multiple microwave signals in parallel. Non-limiting examples of such needs include:
         Simultaneous generation of transmit signal and of a receive local oscillator signal;   Generation of multiple transmit signals in multiple input-multiple output (MIMO) and phased/true delay array systems; and   Generation of sine and cosine local oscillator signals of quadrature down conversion.       

     For example, by digitally modulating the transmit signal and the receive local oscillator signal in a short range frequency-modulated continuous wave (FM-CW) radar system one can introduce an intentional frequency offset so as to avoid handling near-DC signals (see  FIG. 4 ). An inherent trait of this architecture is that several direct conversion blocks and heterodyne converters may be fed from the same synthesizers, thereby naturally meeting the aforementioned need. This allows phase tracking between different microwave signals, as well as tracking of the phase noise. 
     Another advantage of this architecture is the distribution of a generated signal among many nodes, such as transmission antennas/receivers etc. This enables applications such as “Multistatic Radar” (see below). 
     Further embodiments of the present invention provide multiple synthesizers (as in  FIG. 3 ), some of which are modulated and some are not, so as to simultaneously generate multiple signals at arbitrarily spaced frequencies. 
       FIG. 4  illustrates a transceiver according to an embodiment of the present invention. A frequency synthesizer  401  feeds quadrature modulator blocks  403  and  405  having selector switches  411  and  431  respectively, which select between direct output from the quadrature modulator blocks and the outputs of mixers  413  and  433 , respectively, both of which receive input from a frequency synthesizer  407 . The output of selector switch  411  feeds into an amplifier  451 , which in turn feeds an antenna switch/circulator  453  to an antenna  455  for transmission. Signals received from antenna  455  (such as by reflections of the transmitted signal) are fed to a mixer  457 , which receives input from switch  431 . Output of mixer  457  feeds to an anti-aliasing low-pass filter  459  and thence to an analog-to-digital converter  461  (ADC). 
     By modulating quadrature modulation blocks  403  and  405 , fed by the same synthesizer  407  with a frequency shift, both the transmit signal and local oscillator drive for an arbitrary intermediate frequency (IF) receiver are produced. The received signal is down-converted to an intermediate frequency corresponding to the offset of the modulation frequency between quadrature modulation blocks  403  and  405 . 
     Another example of arbitrary waveform modulation-based receiver local oscillator generation is a modulation with a pseudo-random binary sequence (PRBS) modulation, for a spread-spectrum radar. 
     A further example of an arbitrarily-configurable demodulation is multi-tone demodulating. Such a configuration is useful in the simultaneous measurement of several spectral components, e.g. by down-converting them to distinct intermediate frequencies. Both the amplitudes and phases of the spectral components may be measured. 
     The above capability of the signal generator for attaining an output frequency in several configurations, enables relating measurements across the entire frequency range, i.e. including local oscillator and measured path phase. According to a related embodiment, this is achieved by overlapping measurements between different local oscillator frequencies, where the baseband frequencies are adjusted to account for the local oscillator frequency offset between the measurements. This phase-related measurement differs from the common practice in the art, where, as the local oscillator is tuned over the coverage range, unaccounted-for phase changes occur. Retaining the relative phase according to this embodiment is instrumental in characterizing non-linear parameters in a vector network analyzer (VNA) embodiment of the present invention. 
       FIG. 5  illustrates a quadrature receiver according to an embodiment of the present invention. A switch  511  and a switch  531  are ganged together by a common selector  533 , to generate a 0° local oscillator  541  and a 90° local oscillator  543 , which feed mixers  561  and  563 , respectively, to convert a signal received by an antenna  555 , which is amplified by an amplifier  551 . The two intermediate frequency signals are fed into anti-aliasing low-pass filters  571  and  575 , respectively, to be demodulated by analog-to-digital converters  573  and  577 , respectively. 
     The configuration illustrated in  FIG. 5  allows the generation of a 90° split over a wide frequency range, as opposed to conventional analog techniques, and without introducing substantial spurious harmonic content, which occurs when using digital dividers. 
     According to related embodiments of the invention, calibration techniques can be used to adjust the relative phase and amplitude between the quadrature channels. In non-limiting examples: measuring the phase and amplitude between the in-phase (I) and quadrature (Q) components of the down-converted continuous wave signal; simultaneously measuring the phase and amplitude on several signals; and cross-correlation measurements between the I and Q arms. 
       FIG. 6  illustrates a multistatic radar apparatus according to an embodiment of the present invention. In many cases it is desirable for a generated signal to be distributed among many nodes, such as transmission antennas/receivers, and so forth. 
       FIG. 7  illustrates a 3-channel multiple input-multiple output (MIMO) transceiver according to an embodiment of the present invention. In this embodiment, the above-described coherent arbitrary modulation topology is used in conjunction with parallelism (i.e. all quadrature modulation blocks are fed by the same synthesizer and are coherent to each other). This configuration enables active beamforming such as in the context of phased-array antennas. Current implementations are usable principally in narrow-band arrays, where carrier frequencies reach the microwave regime and analog delay-induced phase shifts are used. This embodiment of the present invention provides true beam-forming by digital relative delay means. Beam-forming is achieved by baseband modulation of coherent channels relative to each other, and does not hinder the broad band nature of the transceiver array. In addition, this embodiment provides ease of implementation with digital accuracy. Steering resolution and phase coherence are very precise since the relative phase attainable at any baseband frequency is practically arbitrary, as it is limited principally by digital-to-analog converter resolution. 
     Calibration 
     Calibration plays a significant role, where quadrature modulation imbalance, local oscillator leakage and the response of the receiver and transmitter paths comprise fundamental factors in attaining the required performance of a transceiver. 
     Quadrature modulation imbalance and local oscillator leakage calibrations are typically performed by a minimization of mixing products after passage through a broadband envelope detector. The quadrature modulator is subjected to modulation by complex sine wave at frequency f BB . At the output of the envelope detector, the detected power fluctuates at frequencies associated with the frequency offset between the desired signal and the spurious signals (either 2 f BB  for the quadrature modulation image or f BB  for the local oscillator leakage). The power fluctuations are typically measured by an analog-to-digital converter (ADC). It is important to note that if a high f BB  is used then a high speed ADC is needed in order to capture and quantify the power fluctuations (the ADC bandwidth needs to be at least twice the baseband bandwidth in order to capture both spectral components). 
     Current techniques suffer from inherent difficulties associated with spurious signals and mixing products which fall on the to-be-measured quantities. As an example, mixing products from 2f signal −2f LO  fall on the to-be-measured frequency associated with the quadrature-modulated image: f signal −f image . Thus, the measurements are not independent. Embodiments of the invention facilitate the calibration for quadrature modulation imbalance and local oscillator leakage, without increase in architectural complexity. 
     The corrective action for compensation of quadrature modulation imbalance and local oscillator leakage are well known in the art. The quadrature modulation imbalance compensation involves pre-multiplying the I and Q components by a matrix of correction coefficients. The compensation of local oscillator leakage typically involves adding DC coefficients to the I and Q components. The difficult part of this procedure is determining which coefficients&#39; values to use. This involves a feedback measurement of the strength of the image and spectral components of the local oscillator leakage. 
       FIG. 8  illustrates a spectral component measurement arrangement at the output of the signal generation block according to an embodiment of the present invention. Here, two quadrature modulation blocks are fed by a single, common, synthesizer. A method of measuring the image or local oscillator leakage is by placing the second synthesizer—used to convert the signal to the baseband—at a frequency offset relative to the spectral component of interest. 
     To measure the image, situated at f image =f Sa −f IQa1 , placing the second synthesizer at f Sb =f image −f IF  which will be, after f IF  conversion, linear in the original image magnitude. In order to reach the desired frequency at the output of the second synthesizer—driving the conversion of the output of the quadrature modulation block—fine frequency selection may be facilitated by either/or both the utilization of a fractional N synthesizer and an quadrature modulation of the synthesizer output. Only a single channel (one quadrature modulator, two synthesizers) is needed for the above scheme. 
       FIG. 9  illustrates a receiver-assisted spectral component measurement arrangement according to an embodiment of the present invention. 
       FIG. 10A  illustrates a symmetrized receiver-assisted spectral component measurement arrangement for characterization of a first quadrature modulation block according to the present invention. 
       FIG. 10B  illustrates a symmetrized receiver-assisted spectral component measurement arrangement for characterization of a second quadrature modulation block according to the present invention. 
     Baseband Filter Characterization 
     Baseband filter characteristics may vary at production. In the case of integrated circuit implementation, the filter bandwidth and shape may depend on process, temperature and voltage. The characteristics of baseband filters in the transmit and receive chains may affect system performance regarding signal-to-noise ratio, inter-symbol interference, power flatness, mask conformity, and so forth. It is thus desirable to characterize the filters and compensate for their deviation from desired characteristics. Examples of compensation include directly adjusting the filter and performing digital compensation. 
     The hardware architecture of embodiments of the present invention facilitates measurement of filter characteristics without further increasing complexity. 
     To characterize the transmit filter, the f BB  is scanned throughout the range of interest. For each f BB  the synthesizer&#39;s frequencies (f sa , f sb ) are adjusted such that the resulting intermediate frequency is constant; thus avoiding the receive filter response variation (when measuring at different intermediate frequencies per f BB ). 
     The receiver can be tuned to a frequency corresponding to an aliased frequency±f BB +N·f sample  (where f sample  is the digital-to-analog converter sampling frequency). By doing so, the low pass filter in the transmit path can be characterized beyond the Nyquist frequency of the digital-to-analog converter. 
     Embodiments of the invention as described above and depicted in  FIG. 8  and  FIGS. 10A and 10B  illustrate two similar schemes for scanning the baseband frequency as described above, by digitizing the output of the signal generation block. 
     Measuring the receiver filter is conceptually similar to the above schemes, but benefits from prior knowledge of the transmitter filter response: by knowing the response of the transmission filter, the quadrature modulation frequency can be tuned to scan the frequencies of the receiver filter. Alternatively, it is possible to measure the receiver filter separately without first characterizing the transmission filter. To do so, the quadrature modulation is held at a constant frequency (so as to not incur response variation) and the receiver frequencies are scanned by tuning the synthesizer&#39;s frequencies. 
     The intermediate frequency can be tuned beyond the Nyquist frequency of the analog-to-digital converter so that the receive anti-alias low-pass filter reacts to the input intermediate frequency, while the digitized output is at an aliased frequency±f BB +N·f sample  (where f sample  is the analog-to-digital converter sampling frequency. By doing so, the low pass filter in the receive path can be characterized beyond the Nyquist frequency of the analog-to-digital converter. 
     Self-Characterization of Phase Noise 
     Digitization of the first synthesizer, down converted by the second synthesizer, allows characterizing the relative phase noise between the two synthesizers. This measurement can be used for either self-test purposes or for performance optimizations, such as setting the phase-locked loop parameters so as to optimize the phase noise. An example of such parameter is the setting of the charge pump current in the phase detector. 
       FIG. 11  illustrates a multi-module referenced based scaling arrangement according to an embodiment of the present invention. 
       FIG. 12  is a flowchart  1200  of a method of calibrating a two-synthesizer signal generator according to an embodiment of the present invention. In a step  1201  the first frequency synthesizer is set to the desired test frequency. In a step  1203  an outer loop begins, in which the first numerically-controlled oscillator is set to the desired test frequency offset. In a step  1205 , the second frequency synthesizer and the second numerically-controlled oscillator are set to obtain the desired receiving intermediate frequency. 
     In a step  1207  an inner loop begins for configuring a set of quadrature modulation imbalance correction coefficient values, and in a step  1209  an imbalance-related magnitude is measured. At a decision point  1211 , if the coefficient set is not exhausted, the method returns to step  1207 . Otherwise, if the set is exhausted, the loop beginning in step  1207  exits and the method proceeds to a step  1213 , in which optimal correction coefficients are calculated. 
     At a decision point  1215 , if the first numerically controlled oscillator frequencies are not exhausted, the method returns to step  1203 . Otherwise, if the frequencies are exhausted, the loop beginning in step  1203  exits, and the method concludes with a step  1217 , in which the optimal frequency-dependent correction coefficients are calculated.

Technology Category: 5