Patent Document

This Application Claims Benefit of Provisional Application Ser. No. 60/112785 filed Dec. 18, 1998. 
    
    
     FIELD OF THE INVENTION 
     This invention relates generally to the field of operation amplifiers; and, in particular, to an adaptive biasing scheme for a differential single-pair MOSFET amplifier input stage using backgate biasing techniques to provide a wide common mode range which includes both the positive and negative power supply voltage rails. 
     BACKGROUND OF THE INVENTION 
     Although digital implementations have replaced most analog circuitry, an analog world necessitates the use of operational amplifiers within an integrated circuit to build the interface between external systems and internal electronic circuitry within a variety of electronic devices from medical instruments, portable phones, notebook computers, cassette tape recorders and battery operated electronic devices, to name a few. Operational amplifiers are used primarily with externally applied feedback in pulse shaping, filtering, signal processing and instrumentation applications. In an effort to prolong life and to reduce size and weight of these electronic devices, the industry trend demands a smaller, lower voltage and power consumption operational amplifier. Operational amplifiers can be characterized by their low cost, ease of use and wide availability and, thus, are in high demand. 
     The ideal operational amplifier is a voltage controlled voltage source having a differential input and a single-ended output. Ideal operational amplifier characteristics include infinite gain, zero input offset voltage, infinite input impedance, zero output resistance, high bandwidth, high speed, no frequency dependence, no temperature dependence, no distortion, no processing dependence, sufficient output drive capabilities, and low power consumption. Manufacturing processes, however, generate less than ideal operational amplifier characteristics. Thus, it is the job of the circuit designer to optimize one or more characteristics of the actual operational amplifier in an effort to compensate for non-ideal conditions. 
     Conventional operational amplifier designs include at least two stages: an input stage and an output stage. The input stage, having a non-inverting input and an inverting input, derives the difference between the two inputs. The differential amplifier is one of the most widely used classes of gain stages in analog IC design. As FIG. 1 illustrates, the input stage  10  includes a pair of transistors  12  and  14  configured as differential amplifier having two symmetrical circuit branches, wherein each branch includes a transistor  12  and  14  coupled to one of the input terminals  26  and  28 . Additionally, as active loads each branch includes a second transistor  16  and  18  having directly coupled gates. Each source of transistors  16  and  18  are tied to an upper power supply rail  20 . Each branch beneath differential transistor pair  12  and  14  is coupled to the source of current-source transistor  30 . Current source transistor  30  is biased by a voltage V bias . Within each branch, the transistor produces a signal proportional to the voltage on the corresponding input terminal  26  and  28 . The output  22  and  24  of the input stage  10  is the difference between the signal in each branch  26  and  28  of the differential amplifier. Ideally, the values of corresponding circuit components in the two branches are identical, so that when identical voltages are applied to each input  26  and  28 , i.e. a common-mode input voltage, the signals in each branch are also identical and the output of the input stage  10  is zero. 
     Conventionally, the common-mode input voltage range of a differential gain stage is the maximum range of dc voltage that can be applied, simultaneously, to both inputs without causing the cutoff or saturation of the pair of differential amplifier transistors or the cutoff, saturation, or breakdown of any of the gain stages inside the operational amplifier. A common-mode input voltage which is at or near one of the supply voltages may drive the transistors in the input stage into either a saturation or cutoff condition. This limits the useful range of common-mode input voltages since they must not approach or exceed either of the operational amplifier&#39;s supply voltages. A conventional rule of thumb is that the input signal should not come within about 1 volt of either the high or low power supply rails. 
     In FIG. 1, the lower limit of the input common-mode range is set by the saturation of the current-source transistor  30  having a threshold voltage V T  or the cutoff of gain transistors,  12  and  14 . The lower limit occurs when both inputs are lowered, approaching a voltage within the threshold voltage V T  of the lower power supply rail  34  of voltage−V LL . The upper limit of the common-mode range is set by the saturation of gain transistors,  12  and  14 , as both inputs are raised toward the upper power supply rail  20  of voltage +V HH . Thus, there is normally a high or low-end of the power-supply range, depending upon the polarity of the differential pair transistors  12  and  14 , where the differential pair of transistors  12  and  14  are not operable. 
     Consequently, operational amplifiers of conventional design are limited in range of operable common-mode input voltages. A wide common-mode range, however, is desirable, allowing easy amplifier interface with devices generating input signals at various dc levels. Presently, in single-supply or ground-sensing operational amplifier stages, the range can extend down to the negative power supply rail, −V LL . Yet, there exists no single pair differential amplifier approach that extends the common-mode range to include both the negative and positive power supply rails, +V HH  and −V LL , because the threshold voltage of the differential amplifier pair must be reached prior to each transistor becoming conductive. 
     For this reason, a favored design approach of an input stage within an operational amplifier includes a complementary dual pair of differential amplifiers to compensate for the high or low-end of the power-supply range where one differential pair is operable and the other is not. This complementary dual pair of differential amplifiers has the capability to extend the common-mode range to include both the negative and positive power supply rails, +V HH  and −V LL . Thus, the amplifier is enabled to have rail-to-rail input capability. More particularly, the amplifier output signal represents the differential input voltage as its common-mode portion travels the full extent of the power-supply range. 
     An example of such a design is found in U.S. Pat. No. 5,371,474 which describes several embodiments of a differential amplifier having first and second differential portions operating in parallel to provide representative signal amplification across the full power-supply range. As illustrated in FIG. 2, this input stage  40  having a dual differential amplifier pair  50  and  52  offers a solution to the problems faced with the aforementioned single differential amplifier pair input stage. This proposed approach extends the common-mode range to include both the negative and positive power supply rails. The complementary pair of differential amplifiers  50  and  52  are coupled in parallel such that at least one pair is in operation when the common-mode input voltage is at any voltage within the power-supply range. 
     The first differential amplifier  50  includes a pair of transistors  42  and  44  configured as a differential amplifier having two symmetrical circuit branches, wherein each branch includes a transistor coupled to one of the input terminals  54  and  56 . The second differential amplifier  52  includes a pair of transistors  46  and  48  configured as a differential amplifier having two symmetrical circuit branches, wherein each branch includes a transistor coupled to one of the input terminals  54  and  56 . One of the differential amplifier pairs  50  is active for input signals  54  and  56  at or near upper power rail voltage +V HH , and the other differential amplifier pair  52  is active for input signals at or near lower power rail voltage −V LL . Summing circuit  64  sums the outputs of the two differential amplifier pairs to obtain an output for the input stage  40 . For input signals  54  and  56  that are not near either supply voltage, both of the differential amplifiers are active to a varying degree. Additional circuitry, such as current control circuit  62 , may be incorporated to provide for a smooth transition between states in which only one or the other of the differential amplifiers  50  and  52  is active as a common-mode input voltage varies from one supply voltage  58  to the other  60 . In this way, the common-mode input range is extended to include both power supply voltages  58  and  60 . 
     The first differential portion  50  amplifies a differential input signal by dividing a first tail current I N  into a pair of first main currents, I 1  and I 2 , whose difference is representative of the input signal V 1  when its common-mode voltage V CM  is in the intermediate and high-end ranges. The second differential portion  52  operates in a complementary fashion to amplify the input signal by dividing a second tail current I p  into a pair of second main currents, I 3  and I 4 , whose difference is representative of the input signal when the common-mode voltage V CM  is in the intermediate and low-end ranges. As a result, the differential amplifier has rail-to-rail input capability. 
     However, the increased complexity of an op-amp having dual differential amplifier pairs in the input stage potentially decreases the speed of the amplifier and increases the number of errors during the fabrication process. Just as variations in symmetry between branches of a single differential input stage cause an op-amp of conventional design to exhibit a characteristic input offset voltage, an operational amplifier design employing dual input stage differential amplifiers is subject to similar variations in symmetry between branches of each of its differential amplifiers. 
     Hence, a need exists for a versatile operational amplifier that can be used in a variety of applications powered from battery sources, especially low voltage applications that do not diminish the characteristics of an operational amplifier. A need exists for an operational amplifier input stage that provides high input impedance and a low input offset voltage. A need exists for an operational amplifier that minimizes transistors in the signal path for providing high speed and high bandwidth and still have both input and output rail to rail capabilities. A need exists for a single pair differential gain stage within an operational amplifier capable of a wide common-mode range, inclusive of both negative and positive power supply voltage rails. 
     SUMMARY OF THE INVENTION 
     A single pair differential amplifier gain stage of an operational amplifier having a biasing scheme in accordance with the present invention is capable of providing a wide common-mode voltage range including both positive and negative power supply rails through the use of a biasing scheme which biases the backgate voltages of the differential amplifier transistors. The differential amplifier gain stage circuit is operable between a first and second supply voltage which constitutes a power supply range including a first-end range extending to the first supply voltage and a second-end range extending to the second supply voltage. 
     The bias circuit coupled to the differential amplifier circuit applies a bias voltage to the differential pair of transistors in such a way that the threshold voltages of the differential pair of transistors are adjusted in the response to the common-mode input voltage to turn the differential pair of transistors on when the common-mode input voltage is in a range extending from the first supply voltage to the second supply voltage. 
     Thus, depending on the level of the input common-mode voltage, the biasing scheme alters the threshold voltages of the input gain transistors, enabling a wider common-mode voltage range inclusive of both power supply rails. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     For a more complete understanding of the present invention and the advantages thereof, reference is now made to the following description taken in conjunction with the accompanying drawings in which like reference numbers indicate like features and wherein: 
     FIG. 1 is a schematic of a conventional differential amplifier; 
     FIG. 2 is a schematic of a dual pair differential amplifier in accordance with the prior art having rail-to-rail input capability; and 
     FIG. 3 is a schematic of single pair differential amplifier having rail-to-rail input capability in accordance with the present invention. 
    
    
     DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS 
     FIG. 3 illustrates a schematic of a differential amplifier input stage  90  including the biasing scheme in accordance with the present invention. This input stage  90  provides a wide common-mode voltage range including both positive and negative power supply rails through biasing the backgates of the pair of differential amplifier transistors,  102  and  104 . 
     Specifically, input stage  90  includes a differential amplifier circuit  100  and bias circuit  126 , coupled one to another. The differential amplifier circuit  100  includes a pair of like-polarity differentially coupled field effect transistors,  102  and  104 , configured as a differential amplifier having two symmetrical circuit branches, wherein each branch includes a transistor  102  and  104  coupled to one of the input terminals  114  and  116 , respectively. A pair of like-polarity common-gate field effect transistors,  106  and  108 , are coupled to transistors  102  and  104 , respectively, for providing an active load within each branch. The gates of transistors  106  and  108  are tied together at node  107  where bias voltage V b1  is applied. The source of transistors  106  and  108  are tied to power supply rail  122 . The drain of transistors  106  and  108  are tied to output nodes  110  and  112 , respectively. In addition, the drain of transistors  102  and  104  are coupled to output nodes  110  and  112 , respectively. The pair of like-polarity common-gate coupled field effect transistors,  106  and  108 , are of opposite polarity to the pair like-polarity differentially coupled field effect transistors,  102  and  104 . The drain of a field effect transistor  118  ties each branch together at the source of transistors,  102  and  104 . The source of transistor  118  is connected to the second power supply rail  124 . The gate of transistor  118  is coupled at node  120  to a bias voltage V b2 . 
     Bias circuit  126  includes a replica circuit  130  and a switching circuit  140 , coupled one to another. The replica circuit  130  includes a current source  132 . The drain of a replica field effect transistor  136  is connected in series to the current source  132 . The backgate of replica transistor  136  is coupled to the backgates of the pair of like-polarity differentially coupled field effect transistors,  102  and  104  of differential amplifier circuit  100  for providing backgate voltage to the pair. The replica transistor  136  is equivalent in size to either of the pair of like-polarity differentially coupled field effect transistors,  102  and  104 . An operational amplifier  134  coupled to the drain and backgate of replica transistor  136  supplies a servo loop gain and forms a feedback amplifier to modulate the backgate voltage supplied to the pair of like-polarity differentially coupled field effect transistors,  102  and  104 . The drain of a current source transistor  138  is coupled to the source of replica transistor  136  to supply current equal to one half of the tail current supplied by transistor  118 . Both the gate of replica transistor  136  and the source of the current source transistor  138  is coupled to the second power supply rail  124 . The gate of the current source transistor  138  is coupled to the gate of transistor  118 . The current source transistor  138  is equivalent to half the size the transistor  118 . Thus, replica and current source transistors,  136  and  138  are a replica of half of the input differential stage  100 . 
     The switching circuit  140  includes a field effect transistor  142  coupled in parallel to the series connection of transistors,  136  and  138 , between current source  132  and the lower power supply rail  124 . A field effect transistor  144  is coupled between lower power supply rail  124  and the node that couples the backgates of transistors  102 ,  104  and  136 . The gates of transistors  142  and  144  are coupled to input node  116 . 
     For simplicity, analysis of the circuit is described in the case where the field effect transistors  102 ,  104 ,  118 ,  136 ,  138 ,  142 , and  144 , are n-channel MOS devices and field effect transistors  106  and  108  are p-channel MOS devices. In operation, the differential pair of input transistors,  102  and  104 , amplify a differential input signal  114  and  116  of voltages +V 1 , and −V 1 , by dividing a first tail current I T  into a pair of main currents, I 1  and I 2 . Transistor  118  provides tail current I T  for the pair of like-polarity differentially coupled field effect transistors,  102  and  104 . 
     There are two voltage ranges of operation for the single pair differential amplifier  100 . The first-end voltage ranges from the upper power supply rail voltage +V HH  to voltage V gst , the gate to source voltage V gs  minus the threshold voltage V T  of the transistors  102  and  104 . It is within this range of common-mode voltage V cm  that the transistors  102  and  104  are operable. The second-end voltage ranges from voltage V gst  to the lower power supply rail. It is during the second-end range of input voltage that transistors  102  and  104  are normally cutoff without biasing supplied by bias circuit  126 . To control bias applied in either the first or second-end range, switching circuit  140  switches the replica circuit  130  off and on. When the common-mode voltage V CM  is in a first-end range, the replica circuit  130  is switched off. When the input signal V I  has a common-mode voltage V CM  in the second-end range, switching circuit  140  switches replica circuit  130  on to supply a backgate voltage that adjusts the threshold voltage of transistors  102  and  104 . As a result, the differential pair of input transistors,  102  and  104 , is operable in both first and second-end ranges. 
     Replica circuit  130  is connected to the differential amplifier circuit  100  for replicating half of the input differential stage  100  and supplying backgate voltage to the differential amplifier input transistor pair,  102  and  104 , when necessary. In this manner, the backgates of the differential amplifier input transistor pair  102  and  104  are raised to a voltage higher than the respective source of each transistor  102  and  104 . The assumption is the body effect of each transistor  102  and  104  is sufficiently strong to allow a threshold voltage polarity change without forward biasing the body-source parasitic diode junction. 
     More particularly, during the first-end range of the common-mode voltage V CM , switching circuit  140  including field effect transistors  142  and  144  acts as a switch to turn the biasing of replica circuit  130  off. In this first-end range, the common-mode voltage V CM  exceeds the threshold voltage of transistors  142  and  144 . Consequently, transistors  142  and  144  turn on removing the replica circuit  130  by connecting the drain of replica transistor  136  to the lower power supply rail  124  having voltage −V LL . Since drain of transistor  144  is connected to the backgates of differentially coupled field effect transistors,  102  and  104 , when transistor  144  turns on, the drain to source resistance of transistor  144  becomes very low and effectively the backgates of transistors  102  and  104  are short circuited to the lower power supply voltage −V LL , as well. Accordingly, the body effect is removed since it is not necessary to modulate the back gates of differentially coupled field effect transistors,  102  and  104 . 
     During the second-end range of common-mode voltage V CM , within the replica circuit  130 , current source  132  feeds replica transistor  136 . The magnitude of the current from current source  132  is approximately 80% of that required by the replica current source  138 . Accordingly, the drain voltage of replica transistor  136  rises such that servo loop amplifier  134  pulls up the backgate voltage of replica transistor  136 ; thereby, decreasing the threshold voltage of replica transistor  136  to such an extent that the voltage Vgst of transistor  136  is set to a level which will allow current to pass sent from current source  132  through transistor  136  to transistor  138 . The replica circuit  130  applies this same backgate voltage of replica transistor  136  to the backgates of differential amplifier transistors,  102  and  104 . Thus, when transistors,  102  and  104 , receive an input common-mode voltage V CM  less than V gst  of either transistor  102  and  104  to the voltage of the lower supply rail −V LL , transistors  102  and  104  will pass approximately 80% of the tail current I T  and, thus, are operable. 
     Furthermore, during operation in the second-end range of the common-mode voltage, switching circuit  140  including field effect transistors  142  and  144  acts as a switch which turns the biasing of replica circuit  130  on. In this second-end range, the common-mode voltage V CM  is less than the threshold voltage of transistors  142  and  144 . Consequently, transistors  142  and  144  are not conductive. Thus, the switching circuit  140  does not block the replica circuit  130  from providing backgate biasing to differential amplifier transistors  102  and  104 . In summary, the biasing scheme provides a negative threshold voltage such that the amplifier remains in operation over a wide input common-mode voltage range inclusive of the both power supply rails. 
     As stipulated, the circuit  90  may incorporate field effect transistors  102 ,  104 ,  118 ,  136 ,  138 ,  142 , and  144  as n-channel MOS devices and, consequently, field effect transistors  106  and  108  as p-channel MOS devices. Conversely, the circuit  90  may incorporate field effect transistors  102 ,  104 ,  118 ,  136 ,  138 ,  142 , and  144  as p-channel MOS devices and field effect transistors  106  and  108  as n-channel MOS devices. Thus, in the case where field effect transistors  102 ,  104 ,  118 ,  136 ,  138 ,  142 , and  144  are n-channel MOS devices, the replica circuit  130  provides a backgate voltage which produces a negative threshold voltage. Accordingly, in the case where field effect transistors,  102 ,  104 ,  118 ,  136 ,  138 ,  142 , and  144  are p-channel MOS devices, the replica circuit  130  provides a backgate voltage which produces a positive threshold voltage. 
     Those skilled in the art to which the invention relates will appreciate that various substitutions, modifications and additions can be made to the described embodiments, without departing from the spirit and scope of the invention as defined by the claims. 
     The present invention largely uses FETs. Nonetheless, certain parts of the invention can be alternatively implemented with bipolar transistors. The invention thus can be fabricated in both “CMOS” and “BICMOS” integrated-circuit technologies.

Technology Category: 5