Patent Document

CROSS REFERENCE TO RELATED APPLICATIONS 
   This application claims priority from the provisional application designated Ser. No. 60/306,117 filed Jul. 17, 2001 entitled “High Voltage Generator”. This application is hereby incorporated by reference. 

   BACKGROUND OF THE INVENTION 
   The present invention relates to a power supplies, and in particular to high voltage power supplies. 
   High voltage generator circuits are used to provide power for a variety of applications that rely on the acceleration of charged particles. For example, high voltage generators are used in ion implanter systems for the manufacture of semiconductors, electron beam irradiation systems, x-ray generators, isotope production systems for medicine, research and industry, neutron production systems, accelerator mass spectrometers, research accelerators, and other applications. These applications require the use of high voltage power supplies capable of generating voltages ranging from several kilovolts to a few megavolts, and power levels of several watts to many tens of kilowatts. 
   High voltage generator circuits often include a high voltage multiplier-rectifier circuit, an AC drive circuit, and a transformer interface connecting the AC drive circuit to the multiplier-rectifier circuit. To achieve good performance, it is common practice to operate the AC drive circuit at frequencies ranging from several kilohertz to several hundred kilohertz. It is also common practice to utilize interface transformers with large step-up turns ratios ranging from 10:1 to 1000:1, and multiplier-rectifier circuits comprising a number of cascade stages. 
     FIGS. 1A and 1B  illustrate two known high voltage multiplier circuits. In each of these circuits an AC input signal on a line  20  is coupled to a multiplier-rectifier circuit,  22 ,  24 , respectively, which provides an output signal that is proportional to the input AC voltage amplitude on the line  20  and the number of multiplying stages. To reduce size and cost, the multiplier-rectifier circuits  22 ,  24  typically include many stages and to use capacitors that have the lowest possible capacitance. However, as known, distributed stray shunt capacitance C s  associated with the multiplier-rectifier portion of the power supply limits the performance of multiplier circuits, especially for multiplier circuits employing many stages and low values of coupling capacitance, C c . Alternating currents flowing in the stray shunt capacitance, C s , reduce the output voltage of stages furthest from the AC drive circuit (i.e., from the AC input signal on the line  20 ). 
   The undesirable effects of stray shunt capacitance C s  can be partially overcome by installing a loading inductor, L T , on the last stage as shown in  FIGS. 2A and 2B . Additional loading inductors installed at intermediate locations along the multiplier circuit can further reduce the undesirable effects of stray capacitance. The performance benefits of using loading inductors are disclosed in the publication by E. Everhart, P. Lorain, entitled  The Cockcroft-Walton Voltage Multiplying Circuit , published in the  Review of Scientific Instruments , vol. 24, no. 3, p.221-226, (March 1953). The performance benefits include that the voltage distribution from stage-to-stage can be made substantially more uniform, and the stages furthest from the AC power source can contribute equally or even more than stages close to the AC source. However, the voltage distribution from stage-to-stage becomes dependent on the operating frequency of the AC drive circuit. As disclosed in the above identified publication by Everhart et al., an optimum voltage distribution is defined as one in which the voltage of the first and last multiplier stages, V 1  and V N  respectively, have equal amplitudes. This optimum distribution is obtained when the AC power source is operated at an optimum frequency, ω opt , which depends on the stray capacitance, the coupling capacitance and the loading inductance. 
   The use of the loading inductors also causes the input impedance of the multiplier-rectifier circuit, Z m , to become strongly dependent on frequency. With the addition of loading inductor L T , the multiplier impedance, Z m , exhibits resonant behavior. Below the resonance frequency value the reactive component of multiplier impedance Z m  is inductive, and above resonance frequency value the reactive component of Z m  is capacitive. The optimum voltage distribution as defined above is obtained at a frequency value above the resonant frequency value, and therefore multiplier impedance Z m  has a capacitive reactance when ω=ω opt . At resonance, the ratio of the top stage voltage to first stage voltage, V N /V 1 , achieves its maximum value, and V N /V 1 &gt;1. As the drive frequency is increased above resonance, V N /V 1 , decreases monotonically and is equal to unity at ω=ω opt . Therefore, to achieve good uniformity it is desirable to operate the multiplier-rectifier circuit in the frequency range, ω res &lt;ω&lt;ω opt , which causes the multiplier impedance Z m  to have a capacitive reactance. 
   The interface transformer between the AC drive and the multiplier-rectifier circuit further increases the capacitive load presented to the AC drive circuitry. Interwinding capacitance associated with the secondary winding contributes additional shunt capacitance seen by the AC drive circuit. The winding capacitance appears in parallel with the input terminals of the multiplier-rectifier circuit. 
   It is common practice in the design of high voltage power supplies to achieve efficient coupling by requiring that the AC drive circuit couple to a resonantly tuned circuit, in the case of the multiplier-rectifier circuit described herein, this has been accomplished by incorporating an additional parallel inductor, L IN , at the input terminals of the multiplier-rectifier circuit, or an equivalent inductor in parallel with the primary winding of the transformer, as indicated in  FIGS. 3A-3B , respectively. The inductor value is chosen so that the resonant frequency produced by the inductor and the equivalent capacitance, C EQ , of the multiplier-rectifier circuit, ω=1/√{square root over (L IN C EQ )} is equal to the desired operating frequency, for example ω=ω opt . This approach has drawbacks, especially for commercial applications where ease of servicing and maintenance is important. The system is highly tuned requiring careful adjustment of the driver frequency to the resonant frequency. In addition, a frequency shift of the AC drive or the resonance of the multiplier-rectifier circuit can result from thermal and mechanical effects. Components and subsystems are difficult to replace without retuning of the power supply system. In addition, the input inductor, L IN , is often a source of power loss because of the large circulating currents. 
   Therefore, there is a need for a high voltage power supply system that substantially overcomes the disadvantages of a resonantly tuned AC-drive circuit combined with a multiplier-rectifier circuit, to provide an AC drive circuit that couples power to a load that usually includes a substantial capacitive component. 
   SUMMARY OF THE INVENTION 
   An object of the present invention is to provide a high voltage power supply with substantially uniform voltage distribution along the multiplier-rectifier circuit. 
   Another object of the invention is to provide an efficient technique of driving a multiplier-rectifier circuit that utilizes at least one loading inductor. 
   Yet another object of the invention is to combine a high voltage multiplier-rectifier circuit utilizing a loading inductor, with an efficient non-resonant AC drive that drives the multiplier-rectifier circuit in the frequency range of ω res &lt;ω&lt;ω opt . 
   Briefly, according to an aspect of the invention, a high voltage generator comprises a current source that provides a continuous current signal, which is converted to an alternating current signal by a switching circuit. A multiplier-rectifier circuit with, at least one loading inductor is responsive to the alternating current signal, and provides a rectified output voltage signal. 
   In a preferred high voltage generator, the input of the multiplier-rectifier circuit is briefly short circuited every half cycle of the alternating current signal during the transition from positive to negative current flow. 
   These and other objects, features and advantages of the present invention will become apparent in light of the following detailed description of preferred embodiments thereof, as illustrated in the accompanying drawings. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIGS. 1A and 1B  illustrate known high voltage multiplier circuits; 
       FIGS. 2A and 2B  illustrate known high voltage multiplier circuits that each include a load inductor L T  on their final output stage; 
       FIGS. 3A and 3B  illustrate known high voltage multiplier circuits that each include a load inductor L IN , to resonantly couple the AC-drive circuit to a multiplier-rectifier circuit; 
       FIG. 4A  illustrates a high voltage generator that includes an AC drive circuit configured and arranged to drive a multiplier rectifier circuit; 
       FIG. 4B  is a schematic illustration of an equivalent circuit; 
       FIG. 5A  is a plot of the position of the switches SW 1 , SW 2 , SW 3 , SW 4  and SW 5  ( FIG. 4A ) as a function of time for the operation of the drive circuit of  FIG. 4A ; 
       FIG. 5B  is a plot of the resulting load current waveform and load voltage waveform V 5 -V 6  when the switches are operated according to the timing illustrated in  FIG. 5A ; 
       FIG. 5C  is a plot of the current waveforms for the circuits illustrated in FIG.  4 A and  FIG. 7 ; 
       FIG. 6  illustrates an embodiment of the current source illustrated in  FIG. 4A ; 
       FIG. 7  illustrates an alternative embodiment high voltage generator that includes an AC drive circuit configured and arranged to drive a multiplier rectifier circuit; 
       FIG. 8  is a plot of switch positions for switches SW 1 -SW 4  illustrated in  FIG. 7  as a function of time; 
       FIG. 9  is a schematic illustration of a symmetrical cascade multiplier with two independently adjustable AC-drive circuits; 
       FIG. 10  illustrates a high voltage generator that includes feedback utilizing a voltage divider network; 
       FIG. 11  illustrates high voltage generator that includes feedback utilizing a generating voltmeter; and 
       FIG. 12  is a list of typical values for the components illustrated in FIGS.  4 A- 11 . 
   

   DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 4A  illustrates a high voltage generator  400  that includes an AC drive circuit  402  configured and arranged to drive a multiplier-rectifier circuit  404 . The AC drive circuit  402  includes an adjustable current source  405  that supplies current I on a line  406  to an H-bridge rectifying circuit that includes switches SW 1 , SW 2 , SW 3 , SW 4 ,  408 - 411  respectively. The switches  408 - 411  may be any of the commonly used solid state switching devices such as bipolar junction transistors, insulated gate bipolar junction transistors, metal oxide silicon field effect transistors, or others. The H-bridge converts the continuous current from the current source  405  into an alternating current waveform, which is delivered to primary winding  414  of transformer  416 . Switches SW 1   408  and SW 4   411  operate as a pair, and SW 2   409  and SW 3   410  operate as a pair. The two switch pairs alternately open and close, operating 180 degrees out of phase. When switches SW 1   408 , SW 4   411  are closed, a first lead  420  of the primary winding  414  is connected to a first node  422  (node  3 ) of the current source  405  through switch SW 1   408 , and a second lead  424  of the primary winding  414  is connected to a second node  426  (node  4 ) of the current source  405  through switch SW 4 . When switches SW 2   409 , SW 3   410  are closed, the first lead  420  of the primary winding  416  is connected to the second node  426  (node  4 ) of the current source, and the second lead  414  of the primary winding  416  is connected to the first node  422  of the current source  405 . 
     FIG. 4B  is a schematic illustration of an equivalent RLC circuit of the transformer  416  and the voltage multiplier-rectifier circuit  404 , representing the impedance in the frequency range of interest (i.e., ω res &gt;ω&gt;ω opt ). One of ordinary skill in the art recognize that the components illustrated in  FIG. 4B  are not actual physical components, but rather represent equivalent circuit values of the transformer  416  and multiplier-rectifier circuit  404 . Capacitor C 1    440  is the transformed impedance of the multiplier-rectifier circuit shown in  FIG. 4A  when operated at a frequency close to the optimum frequency, discussed above. Inductor L 1    442  is the equivalent series load inductance including the leakage inductance of the transformer  416  (FIG.  4 B), the inductance of interconnecting cables (now shown), and any additional lumped or distributed inductance between the AC drive circuit  402  and the multiplier-rectifier circuit  404 . Resistor R 1    444  is the resistive load transformed to the primary winding  414  (FIG.  4 A). 
   A difficulty encountered when driving loads that have large capacitive reactances, are the high drive currents required to repetitively reverse the polarity of the load voltage. This is especially true at high frequency and for non-sinusoidal waveforms encountered with switching converters. These high currents in combination with phase shifted voltage and current waveforms place demands on the output switching devices, and result in power losses. The drive circuit shown in  FIG. 4A  overcomes these difficulties by incorporation of a shorting switch, SW 5   430 . 
     FIG. 5A  is a plot of the position of the switches SW 1 , SW 2 , SW 3 , SW 4  and SW 5  ( FIG. 4A ) as a function of time for the operation of the drive circuit of FIG.  4 A.  FIG. 5B  is a plot of the resulting load current waveform and load voltage waveform V 5 -V 6 . Referring to  FIGS. 4A ,  5 A, and  5 B, the switches SW 1 -SW 4  ( FIG. 4A ) operate as an H-bridge circuit alternately reversing the direction of current flow through the load. The switch SW 5   430  ( FIG. 4A ) conducts for a short period of time during the transition from the switch pair SW 1 , SW 4  to the switch pair SW 2 , SW 3 . The switch SW 5   430  is closed for a sufficient time period to cause current reversal to take place in the transformer  416 . This relies on the resonant interaction of equivalent components L 1  and C 1  shown in the RLC equivalent circuit of FIG.  4 B. The switch SW 5  must not remain closed for too long a time period to allow oscillation of the load current. Preferably, the switch SW 5   430  remains closed for a time period approximately equal to one quarter of the oscillation period, T SW5 ≈(2/π)√{square root over (L 1 C 1 )}. 
   Referring again to  FIG. 4A , notably the H-bridge is driven with a current source  405  rather than the common practice of using a voltage source. During the time period when the load shorting switch SW 5   430  is closed, the voltage between nodes  422 ,  426  is about zero (i.e., V 7 -V 8  ≈0), and a continuous current flows through the H-bridge switches. The current source  405  does not deliver power to the circuit during this time period, except for the power produced by the small voltage drop present in non-ideal switches. The current source  405  is not called upon to provide or store energy for the purpose of reversing the output polarity of the AC-drive circuit  402 . Energy stored in the equivalent load capacitance C 1 , as shown for convenience in  FIG. 4B , during the positive (or negative) polarity of the waveform is recovered and used to reverse the polarity of the load current and voltage. An additional benefit of using a current source  405  to drive the H-bridge is the inherent protection afforded to the H-bridge switches SW 1 -SW 4 . If unintended switching of the switching devices or accidental shorting of output node  5   460  and output node  6   462  takes place, the current flowing in the switching devices is limited by the current from the current source  405 . This is in contrast with a voltage driven H-bridge where either of these fault conditions inevitably results in failure of one of the switching devices.  FIG. 5C  is a plot of the current waveforms for the circuits illustrated in FIG.  4 A and FIG.  7 . 
     FIG. 6  illustrates an embodiment of the current source  405 , employing a forward converter current source. Although this is not an ideal current source, it provides the benefits discussed above. Current flow is maintained at a relatively constant value with the use of a series inductor L 2   602 . The operation of this circuit is well known. Switch SW 6   604  alternately opens and closes injecting current into inductor L 2   602 . During the time period when the switch SW 6   604  is closed, the inductor L 2   602  is connected to voltage source V 1   608 , and the current in the inductor L 2   602  increases at a rate dI L2 /dt=V L2 /L 2 . While the switch SW 6   604  is open, the continuous current of the inductor L 2   602  flows through free wheeling diode D 5   610 . The rate of rise in the current delivered to the H-bridge during the time period when the switch SW 5  is closed depends on the value of inductance L 2   602  and the relative timing of the switch SW 6   604 . 
     FIG. 7  illustrates an alternative embodiment high voltage generator  700  that includes an AC drive circuit configured and arranged to drive a multiplier rectifier circuit. This high voltage generator  700  is substantially the same as the high voltage generator illustrated in  FIG. 4A , with the principal exception that the voltage generator does not include a shorting switch SW 5 . Specifically, the high voltage generator  700  does not use the switch SW 5   430  ( FIG. 4A ) to momentarily short the output nodes  5  and  6  of the H-bridge. 
     FIG. 8  is a plot of switch positions for switches SW 1 -SW 4   702 - 705  (FIG.  7 ), respectively illustrated in FIG.  7 . Significantly, output shorting is accomplished by causing the two switch pairs SW 1 , SW 4  and SW 2 , SW 3  to be simultaneously closed for a short time period during polarity reversal of the waveform, thus performing the same function as switch SW 5   430  (FIG.  4 A). 
   The inventive high voltage power supply incorporates the AC-drive circuit depicted in  FIG. 4A  with various multiplier-rectifier circuits to provide an efficient high voltage generator. In particular, multiplier-rectifier circuits which have been optimized for voltage uniformity by utilizing one or more loading inductors, are well suited for the inventive generator. These multiplier-rectifier circuits, as well as others, exhibit an input impedance that is substantially capacitive. This is especially true for multiplier-rectifier circuits that have been optimized for voltage uniformity, minimum stored energy, or highest average voltage per cascade stage. When combined with the AC-drive circuit of  FIG. 4A , the resulting high voltage generator obviates the need for resonantly tuning the AC-drive circuit to the multiplier-rectifier circuit. The performance and efficiency of the high voltage generator is relatively insensitive to the operating frequency. In addition, individual components or subassemblies can be replaced or interchanged without the need for retuning the power supply system. 
     FIG. 9  depicts another embodiment of a high voltage generator  900 . The voltage generator  900  includes a symmetrical multiplier-rectifier circuit  902  utilizing a loading inductor  904 , two interface transformers  906 ,  908 , and two current driven H-bridge AC-drive circuits  910 ,  912 . The AC-drive circuits are operated 180 degrees out of phase, and individually provide power to the two symmetrical legs of the multiplier-rectifier circuit. The amplitude of the two AC-drive circuits may be individually adjusted to obtain optimum balance in the multiplier-rectifier circuit. Balance is achieved when minimum voltage ripple on the high voltage terminal of the generator is attained. The balance adjustment allows for compensation of variations in component values, tolerances and mechanical assembly. 
   In the inventive high voltage generator, the DC output voltage amplitude of the generator is controlled by adjusting the amplitude of the steady state current source of the AC-drive circuit. Stabilization of the high voltage output may be accomplished by the use of feedback. Compensation for variations in high voltage output is accomplished by changing the output of the AC-drive in response to variations in high voltage output. The high voltage output may be monitored using a voltage divider network that includes resistors or resistor and capacitors. The measured voltage in combination with a feedback amplifier and compensation circuit is used to adjust the amplitude of the current from the current source to compensate for variations in output voltage, as illustrated in FIG.  10 . 
   Referring to  FIG. 11 , feedback stabilization may also be accomplished by measuring the high voltage output with a generating voltmeter (GVM)  1102 . The GVM  1102  may be a rotating vane GVM, or a vibrating capacitor GVM. Using a GVM for feedback stabilization has several advantages when compared to the high voltage divider technique. The GVM is a non-contacting measurement technique and does not draw current from the high voltage generator. In addition, the GVM is less susceptible to inaccuracies due to thermal and voltage coefficient effects, and the GVM exhibits relatively fast response times. Furthermore, the GVM is usually less affected by stray current paths such as corona and stray capacitance. 
     FIG. 12  is a list of typical values for the components illustrated in  FIGS. 4A-11 . 
   A preferred embodiment of the inventive high voltage generator incorporates feedback stabilization preferably using a GVM voltage measuring technique. 
   Although the present invention has been shown and described with respect to several preferred embodiments thereof, various changes, omissions and additions to the form and detail thereof, may be made therein, without departing from the spirit and scope of the invention.

Technology Category: 5