Patent Abstract:
A photovoltaic module-mounted AC inverter circuit uses one or more integrated circuits, several switches, solid dielectric capacitors for filtering and energy storage, inductors for power conversion and ancillary components to support the above elements in operation. The integrated circuit includes all monitoring, control, and communications circuitry needed to operate the inverter. The integrated circuit controls the switches in both an input boost converter and a single-phase or multi-phase output buck converter. The integrated circuit also monitors all power processing voltages and currents of the inverter and can take appropriate action to limit power dissipation in the inverter, maximize the available power from the associated PV module and shut down the inverter output if the grid conditions so warrant. The integrated circuit implements power line communications by monitoring the AC wiring for signals and generating communications signals via the same pulse-width modulation system used to generate the AC power.

Full Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
       [0001]    This application is a continuation of co-pending U.S. non-provisional patent application Ser. No. 12/121,580 filed May 15, 2008, which further claims the benefit of U.S. provisional patent application Ser. No. 60/938,663 filed May 17, 2007, both of which are herein incorporated by reference in their entirety. The present application is further related to co-pending U.S. patent application Ser. No. 12/121,578 entitled, “Photovoltaic AC Inverter Mount and Interconnect” and filed May 15, 2008, and U.S. patent application Ser. No. 12/121,616 entitled, “Distributed Inverter and Intelligent Gateway”, and filed May 15, 2008, both of which are hereby incorporated by this reference in their entirety. 
     
    
     BACKGROUND OF THE INVENTION 
       [0002]    1. Field of the Invention 
         [0003]    The invention relates to direct current to alternating current power conversion and, more particularly, to photovoltaic module output power conversion to alternating current. 
         [0004]    2. Description of the Related Art 
         [0005]    An inverter is a device that performs direct current (DC) to alternating current (AC) power conversion.  FIG. 1  shows a functional diagram of a prior art inverter  101  implementing the power conversion process, which is described in further detail below. 
         [0006]    Inverters can be designed to supply power from photovoltaic (PV) modules to the utility power grid, otherwise simply known as the grid. The process of supplying power to the grid places several special constraints on the power conversion process. First, there exists an optimum voltage across the PV module terminals at which maximum power is to be extracted. This is denoted as the maximum power point and it is found via various measurements and computational algorithms. Second, the utility power grid signal appears as a voltage source with low impedance. The best drive signal from the inverter into the utility power grid is a current. Third, the inverter AC output current must be synchronous with the utility power grid voltage. If it is not synchronous, a non-unity power factor may exist resulting in the transfer of undesirable reactive power or, in an extreme case, no power is efficiently transferred from the inverter into the utility power grid due to a significant frequency or phase difference. Fourth, the inverter must monitor the utility power grid and, if there is a failure of the grid supply, prevent any current flow from the inverter into the grid. Grid failure may be due to a break in the grid wiring to the inverter site. Under this condition, if the inverter drives the grid, the remaining portion of the grid connected to the inverter is energized. Since a limited region of the disconnected grid is now energized, it becomes an island of power relative to the dimensions of the grid. Prevention of the island condition by the inverter grid detection mechanism is known as anti-islanding. Anti-islanding is important in that utility workers can be exposed to the hazard of undesired power in an island and have no means to reliably determine if an island exists or disable the power entering the island, particularly if the grid problem is physically distant from the inverter driving the grid. 
         [0007]    Existing photovoltaic inverters generally fall into the category of centralized inverters. The centralized inverter accumulates DC power from multiple PV modules wired in series or series combined with parallel connections to achieve a significant total power. This power is converted to AC within the centralized inverter and is connected to the grid. The expected benefit of this method is that the high DC voltage of a series connected string of PV modules allows for greater efficiency in power conversion. Another benefit is that control and monitoring of the system is also centralized. 
         [0008]    There is also a category of distributed inverters in which multiple inverters are used to generate the desired AC power from a number of PV modules. In an extreme case, one inverter can be assigned to convert power from one PV module. If the inverter is mounted on the PV module, the assembly comprising the PV module and inverter is termed an AC module. AC modules are generally connected in parallel as opposed to the series connection typically seen for multiple DC connected PV modules used with a centralized inverter. 
         [0009]    The benefits of AC modules are multi-fold. First, if an inverter or PV module fails, all other modules can still provide their full power capacity resulting in minimal impact on the total power produced by a PV system. Second, effects of shading or other means that cause one PV module to operate at reduced current does not affect the operation of other modules. In a centralized system, the series connection means that the PV module with the lowest output current limits the entire string of PV modules to this current, regardless of illumination conditions resulting in an overall loss of power on the order of 10 to 30 percent under typical conditions. Third, the inverter in the AC module is capable of measuring the power output of its associated PV module and, via communications means, can report this data to external devices. Centralized inverter systems require an additional, relatively expensive, sensing and communications system to be mounted at each PV module to be able to monitor the performance of individual PV modules. Other advantages have been documented in the prior art. 
         [0010]    Previous attempts at development and marketing of AC modules have met with little or no success. The primary reason has been that the sales volume was too low to achieve any kind of economy of scale. The components used in the associated inverters were off-the-shelf, and in many cases were not optimum for the application. The inverter lifetime was limited by many of these components, especially the electrolytic capacitors used for energy storage. The reliability levels of existing off-the-shelf components used in the inverters has limited their lifetime to between five and ten years. 
         [0011]    What is desired, therefore, is a circuit that uses a minimum number of components, uses no limited-reliability components and has been optimized for large-scale manufacturing. Such a device would simultaneously achieve economy of scale to support the rapid adoption of solar power via PV modules and can remain in operation for a period of time consistent with other structural electrical power systems. 
       SUMMARY OF THE INVENTION 
       [0012]    According to the present invention, a PV module-mounted AC inverter circuit uses one or more integrated circuits, several power transistors configured as switches, several solid-dielectric capacitors for filtering and energy storage, several inductors for power conversion and ancillary components to support the above elements in operation. 
         [0013]    The integrated circuit is developed to include all monitoring, control and communications circuitry needed to operate the inverter. In particular, it controls the activity of pulse-width modulated power handling transistors in both an input boost converter and a single-phase or multi-phase output buck converter. A high voltage bus connects the two converters and has charge storage capacitors to maintain the high voltage output of the boost converter. The integrated circuit also monitors all power processing voltages and currents of the inverter and can take appropriate action to limit power dissipation in the inverter, maximize the available power from the associated PV module and shut down the inverter output if the grid conditions so warrant. The integrated circuit implements power line communications by monitoring the AC wiring for signals and generating communications signals via the same pulse-width modulation system used to generate the AC power. Communications is used to report inverter and PV module status information, local identification code and to allow for remote control of inverter operation. 
         [0014]    A single-inductor, grounded-input, bipolar-output boost converter is used to convert the relatively low voltage of the PV module to a higher voltage suitable to drive the output buck converter. The boost converter has two operating modes allowing for optimization of the inverter system design. 
         [0015]    A method of parallel connecting two single-inductor, bipolar-output boost converters can also be used to reduce the current requirements of the inductors, transistor switches and diodes of a single boost converter while maintaining the same output power as for the single boost converter. The two converters operate in quadrature phase so that neither inductor is simultaneously discharged. A further benefit of this connection is an increase in ripple frequency present at the boost converter inputs and outputs that allows for a reduction in ripple filtering capacitor value, size and cost. More boost converters can be connected in parallel resulting in further overall improvements. 
         [0016]    A method of utilizing delta-sigma modulation in generating the pulse-width modulated drive to the power transistors is given. A standard digital pulse-width modulator with low clock frequency and commensurately low timing resolution is used. A delta-sigma modulator is then used to dither the pulse width up or down by a single clock cycle on a pulse-by-pulse basis. The spectral shaping of the delta-sigma modulation then improves the low frequency resolution of the pulse-width modulated (PWM) system such that a much higher averaged equivalent pulse-width resolution, easily up to millions of times improved, is possible without increasing the PWM clock or PWM output frequency. 
         [0017]    A method of input and output current measurement for the input boost converter utilizing the inductor voltage and timing signals is given. The inductor voltage is measured and converted to an equivalent digital value via an analog-to-digital converter (ADC). The single-inductor boost converter switch pulse widths are multiplied by the inductor voltage, averaged over a full inductor switching cycle period and divided by the known inductance to calculate the input current. The time the switches are disabled is multiplied by the inductor voltage, averaged over a full inductor switching cycle period and divided by the known inductance to calculate the output current. 
         [0018]    A method output current measurement for the output buck converter utilizing the inductor voltage is given. The inductor voltage is measured and converted to an equivalent digital value via an ADC. The digital value is passed through a low-pass digital filter to extract a short-term average sufficiently fast to allow for precise current amplitude and phase measurement at the power line frequency. The resulting digital value is divided by the known inductor DC resistance to determine the value of current at the AC output of the inverter. This process is performed for each output phase of a multi-phase output inverter. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0019]    The present invention is illustrated by way of example and not by limitation in the accompanying figures in which like reference numerals indicate similar elements and in which: 
           [0020]      FIG. 1  is a block diagram of a prior art PV inverter including a boost converter, high voltage DC bus, charge storage, buck converter, monitoring system and control system; 
           [0021]      FIG. 2  is a block diagram of the present invention PV inverter including a boost converter, high voltage DC bus, charge storage, buck converter and an integrated circuit comprising a monitoring system, control system and communications system; 
           [0022]      FIG. 3  is a block diagram of the present invention PV inverter integrated circuit; 
           [0023]      FIG. 4  is a schematic of the present invention single-inductor, bipolar-output boost converter; 
           [0024]      FIG. 5  is a timing diagram depicting the timing sequence of signals occurring in the present invention bipolar-output boost converter using a single inductor operating in two-cycle mode; 
           [0025]      FIG. 6  is a timing diagram depicting the timing sequence of signals occurring in the present invention bipolar-output boost converter using a single inductor operating in single-cycle mode; 
           [0026]      FIG. 7  is a schematic of the present invention tandem single-inductor, bipolar-output boost converter; 
           [0027]      FIG. 8  is a timing diagram depicting the timing sequence of signals occurring in the present invention tandem bipolar-output boost converter using a single inductor operating in single-cycle mode; 
           [0028]      FIG. 9  is a schematic of the present invention boost converter inductor current indirect measurement system; 
           [0029]      FIG. 10  is a schematic of a prior art multi-phase output buck converter; 
           [0030]      FIG. 11  is a schematic of the present invention buck converter inductor current measurement system; 
           [0031]      FIG. 12  is a block diagram of a prior art digital pulse-width modulator; and 
           [0032]      FIG. 13  is a block diagram of a pulse-width modulator utilizing sigma-delta modulation to simulate fine resolution adjustment of the pulse width. 
       
    
    
     DETAILED DESCRIPTION 
       [0033]    Referring now to  FIG. 1 , a PV module-mounted inverter block diagram  101  according to the prior art includes a DC input  104  for receiving direct current power from a PV module. The current is converted from the DC input  104  to a higher voltage by the boost converter  105  to drive the high voltage DC bus  107 . The generated voltage is maintained on the high voltage DC bus  107  by charge storage capacitors  106 . The high voltage is converted to an AC output  109  by buck converter  108 . The AC signal is compatible in frequency and phase to the AC load signal provided at AC output  109 . Monitoring system  102  detects the relevant input voltage and current at DC input  104 , the voltage at the high voltage DC bus  107  and the output voltage and current at AC output  109 . The detected signals are converted by monitoring system  102  into a suitable format for processing by control system  103 . Control system  103  generates appropriate signals to operate and control the boost converter  105  and buck converter  108 . A first capability of the monitoring system  102  is to determine the required DC input  104  voltage to extract maximum power from the associated PV module. A second capability of monitoring system  102  is to detect the voltage and frequency of the signal at AC output  109  to determine if the AC output voltage and frequency are within predetermined limits of operation for the inverter. If the limits are exceeded, the buck converter  108  is disabled by the control system. A third capability of the monitoring system is to compare the phase of the AC output voltage to the AC output current at AC output  109 . The phase must be maintained to cause power to flow out of the AC output  109  with a power factor substantially close to unity. A first capability of control system  103  is to generate timing signals to cause boost converter  105  to convert the low DC input  104  voltage to the high voltage at high voltage DC bus  107 . A second capability of control system  103  is to generate timing signals to cause buck converter  108  to convert the high voltage at high voltage DC bus  107  to an AC output current at AC output  109 . The prior art PV panel-mounted inverter  101  does not report data directly to any other system. The prior art PV module-mounted inverter  101  utilizes between 200 and 1000 standard components to perform its functions. The large number of components requires a large dimension for the enclosure of the inverter and significantly reduces the reliability and maximum lifetime of the device. 
         [0034]    Referring now to  FIG. 2 , a PV module-mounted inverter block diagram  201  according to the preferred embodiment of the present invention includes a DC input  202  and a boost converter  203  to convert the DC input  202  voltage to a high voltage at high voltage DC bus  205 . The voltage is maintained by charge storage  204  to minimize impact of loading. Buck converter  206  converts the high voltage to a current at AC output  207 . Integrated circuit  210  contains monitoring system  211 , communications system  212  and control system  213 . Monitoring system  211  detects voltage and current at DC input  202 , voltage at high voltage DC bus  205  and voltage and current at AC output  207 . The detected signals are converted by monitoring system  211  into a suitable format for processing by control system  213 . Control system  213  generates appropriate signals to operate and control the boost converter  203  and buck converter  206 . A first capability of the monitoring system  211  is to determine the required DC input  202  voltage to extract maximum power from the associated PV module. A second capability of monitoring system  211  is to detect the voltage and frequency of the signal at AC output  207  to determine if the AC output voltage and frequency are within predetermined limits of operation for the inverter. If the limits are exceeded, the buck converter  206  is disabled by the control system. A third capability of the monitoring system is to compare the phase of the AC output voltage to the AC output current at AC output  207 . The phase must be maintained to cause power to flow out of the AC output  207  with a power factor substantially close to unity. A fourth capability of the monitoring system is to determine if the inverter has been disconnected or connected to the PV module or AC grid. If a disconnection occurs, the monitoring system detects the sudden change in terminal signals and sends a signal to the control system to shut down the inverter to minimize contact arcing while being disconnected. During the applications of connections, the monitoring system establishes that conditions are met for proper operation with all connections secure for a minimum time period prior to enabling the inverter to eliminate any arcs during the application of connections to the inverter. This capability is termed “hot-swap” and minimizes the difficulty in installation or replacement of an inverter. A first capability of control system  213  is to generate timing signals to cause boost converter  203  to convert the low DC input  202  voltage to the high voltage at high voltage DC bus  205 . A second capability of control system  213  is to generate timing signals to cause buck converter  206  to convert the high voltage at high voltage DC bus  205  to an AC output current at AC output  207 . Communications system  212  provides a means for external communications with the PV panel-mounted inverter  201  via the AC output  207  utilizing power line communications. Data to be sent to the inverter  201  is encoded via carrier signal on the AC load voltage by an external communications device and appears at AC output  207 . Monitoring system  211  detects this signal and converts it from an analog signal into a digital signal suitable for processing by communications system  212 . Communications system  212  maintains a local address unique to each inverter in a system and detects the existence of a message for the present inverter  201 . If the message address does not match the local address, no action is taken by the communications system  212 . If the message address matches the local address, then communications system  212  determines a control action depending on the message content. Control actions may include, but are not limited to, enabling or disabling the inverter, adjusting inverter performance parameters such as output voltage, current or power, changing maximum power-point tracking operating modes and initiating inverter  201  test functions. Data to be sent by inverter  201  to external devices is encoded by communications system  212  onto a carrier signal to be applied to AC output  207 . The carrier could be generated in inverter  201  or it could load modulate an externally generated carrier arriving at AC output  207 . Communications system  212  sends the encoded data to control system  213  to generate pulses that cause buck converter  206  to either generate a modulated carrier signal or to load modulate an external carrier signal at AC output  207 . In this way, buck converter  206  is used for both generation of AC line power and carrier communications signals at AC output  207 . The present invention PV module-mounted inverter  201  thus supports power line communications with external devices via AC output  207  and removes the typical requirement for an additional communications medium, such as independent wires or wireless radio frequency, to support external communications with the inverter  201 . The assembly of the monitoring system  211 , communications system  212  and control system  213  into integrated circuit  210  results in a total component count for the present invention PV module-mounted inverter  201  of between 20 and 80 components. This is significantly less than that for the prior art PV module-mounted inverter  101  and leads to a significant reduction of present invention PV module-mounted inverter  201  cost and a tremendous improvement in reliability. Reliability is further enhanced since a custom integrated circuit  210  can be designed to achieve a lifetime of 25 years that well exceeds the lifetime of 10 years for typical off-the-shelf commercial integrated circuits. The integrated circuit  210  may also include active and passive components of boost converter  203  and buck converter  206 . Integrated circuit  210  may also be implemented as multiple integrated circuits. Charge storage  204  is normally implemented as a capacitor, but may also be implemented as flux storage by utilizing an inductor and changing DC bus  205  to a current mode of operation. A person skilled in the art can also combine the functions of the boost converter, charge storage and buck converter via circuit manipulations. Monitoring system inputs can be voltages, currents, voltages that are representative of circuit voltages and currents or currents that are representative of circuit voltages and currents. The various signals in the inverter can be implemented as analog, sampled analog or digital signals. Digital signals can be implemented in parallel word form or serial bit-stream form. 
         [0035]    Referring now to  FIG. 3 , a PV module-mounted inverter integrated circuit block diagram  301  according to a preferred embodiment of the present invention includes a DC input and internal monitor inputs  302 ; power supply and regulator  303 ; waveform generator, sync and amplitude detect  304 ; current and voltage sensors and ADC  305 ; power transistor drive outputs  306 ; transistor control pulse generator, level shifter and delta-sigma modulator  307 ; error analyzer and loop stabilization  308 ; communications interface  309 ; controller  310 ; identification (ID) code store  311  and AC output and AC monitor inputs  312 . Power supply and regulator  303  converts the grid voltage arriving at AC output  312  into a DC voltage to bias all of the integrated circuit  301  blocks, boost converter  203  transistor controls and buck converter  206  transistor controls via rectification and filtering. The total power utilized from the power supply and regulator  303  is very small allowing for inclusion in integrated circuit  301 . Current and voltage sensors and delta-sigma modulator ADC  305  detect all input voltages appearing at DC input and internal monitor inputs  302  and output and AC monitor inputs  312  and convert them to an equivalent digital code via an ADC. This code is transmitted to other blocks to establish the operating conditions for control of the inverter  201  and to report the operating conditions via power line communications. Waveform generator, sync and amplitude detect  304  examines the AC monitor input  312  to determine the frequency, phase and amplitude of the AC grid voltage. It generates a digitally-encoded sine wave that is synchronized to the AC grid voltage with as many phases as appear at AC monitor input  312 . Error analyzer and loop stabilization  308  compares the sine wave generated by the waveform generator  304  with the AC output current measured via AC monitor inputs  312  and the current sensor  305  and generates an error feedback signal. The error feedback signal is passed through a dynamic, reconfigurable feedback loop stabilization filter containing an integrator and is fed to pulse generator  307 . Transistor control pulse generator, level shifter and delta-sigma modulator  307  converts the filtered digital error signal word into a pulse waveform. The pulse waveform can be a simple digitally generated PWM signal, a delta-sigma modulated fixed pulse-width signal or a PWM signal with delta-sigma dithering to achieve finer resolution than the simple digitally generated PWM alone. The generated pulse is fed to a level shifter that converts the low voltage logic signals of the internal integrated circuit logic to high voltage signals suitable to drive the boost converter  203  and buck converter  206  power processing transistor control inputs. The level shifter limits the drive voltage or current to the power processing transistors to prevent damage to those devices. Controller  310  manages the operation of integrated circuit  301  and inverter  201 . Controller  310  continuously monitors the sensors and ADC  305  outputs to determine the state of the DC PV module power at the DC input  302 , the grid voltage and the current into the grid at the AC output  312 . If the grid voltage amplitude and frequency fall out of the allowed inverter operating range, the controller disables the output so as to avoid the grid islanding condition. If the grid voltage is within the allowed amplitude and frequency specifications, the controller checks the communications interface  309  to determine if there are any communications conditions to be applied to the inverter activity. If the communications channel is available and a signal is sent commanding the inverter to be enabled, the controller  310  will start operation of the pulse generator  30  and error analyzer and loop stabilization  308 . Otherwise, these circuits are disabled and no AC output current flows from the inverter. Communications of control and status information is routed through communications interface  309 . The controller  310  receives commands from and can transmit commands to the communications interface  309  to interact with external circuits via power line communications. Commands to the inverter would include, but not be limited to, enabling or disabling the inverter and requests for inverter data or initiation of an inverter self-test. Data from the inverter would include, but not be limited to, all available input and output current and voltage information, self-test results and inverter identification code number. The inverter identification code number is stored in ID Code Store  311 . This is a large digital number that uniquely identifies the inverter for purposes of communications and tracking. The identification code is used during communications as an address for an external communications device to specifically select the inverter for commands and responses. The identification code is also used to track the device during manufacture, distribution and installation to assist in determination of inventory, security, failure analysis and for other related purposes. A person skilled in the art could implement the functions for integrated circuit  301  in multiple integrated circuits or as a combination of integrated circuits and discrete components. The controller  310  can be implemented using state machines, microprocessor or microcontroller techniques known in the art. The ID code store  311  may be implemented using fuses, printed circuit layout elements, programmable read-only memory, flash memory, ferroelectric memory, fuse memory or other such means of data storage as exists in the field of non-volatile electronics data storage. 
         [0036]    Referring now to  FIG. 4 , a single-inductor, bipolar-output boost converter  401 , according to an embodiment of the present invention, includes a DC input  402 , an input switch PWMI  403 , an input switch current ISW  412 , a ground switch PWMG  404 , a negative output diode  404 , a negative output diode current  413 , a negative output DC voltage  405 , a negative output capacitor  407 , an inductor  406 , a positive output diode  409 , a positive output diode current  414 , a positive output DC voltage  410 , a positive output capacitor  411 , an inductor current IL  412  and a ground terminal  413 . The switches  403  and  408  are closed in a timing sequence to implement one of two methods of generating a simultaneous positive  410  and negative  405  output voltage from a positive input voltage  402  referenced to ground  413 . Closing both PWMI  403  and PWMG  408  causes a current IL to flow in inductor  406 . The current rises in a linear slope during the inductor  406  charge cycle. After allowing the current to rise to a maximum value, one of the switches  403  or  408  is opened. Opening PWMG  408  causes the inductor current IL to flow into diode  409  and then into capacitor  411 . The capacitor voltage is charged in a positive direction relative to ground  413 . If instead switch PWMI  403  is opened, the inductor current  406  will flow through diode  404  and from capacitor  407 . The capacitor voltage is charged in a negative direction relative to ground  413 . In the first mode of operation, the generation of positive and negative currents is performed by charging the inductor  406  and discharging it into capacitor  411  resulting in a positive voltage at output  410 . Then the inductor  406  is charged again by closing both switches  403  and  408  and it is discharged into capacitor  407  resulting in a negative voltage at output  405 . The total sequence of two charge and discharge cycles of inductor  406  is repeated resulting in a continuous positive  410  and negative  405  output voltage from the boost converter. Thus this first mode is termed the two-cycle mode of operation. In the second and preferred mode of operation, inductor  406  is charged by closing both switches  403  and  408 . Switch PWMI  403  is then opened resulting in current in diode  404  from capacitor  407 . A short time later, switch PWMG  408  is opened and the current now flows in both diode  404  and diode  409 . A positive voltage results on output  410  and a negative voltage results on output  405 . This sequence repeats resulting in a continuous positive  410  and negative  405  output voltage from the boost converter. Since the inductor  406  is charged and discharged only once to create both positive and negative output voltages simultaneously, this second mode is termed the single-cycle mode of operation. Alternative input connections to the boost converter  401  can be implemented by a person skilled in the art by connecting switch PWMG  408  to a negative input source and switch PWMI  403  to ground  413  resulting in a single inductor converter utilizing a negative input voltage. First diode  404  and second diode  409  could be replaced with switches to implement a synchronous rectification for both positive  410  and negative  405 . Relative timing of the switches and diode currents can be adjusted by one skilled in the art to optimize overall system efficiency without changing the basic operation of the boost converter  401 . 
         [0037]    Referring now to  FIG. 5 , a timing diagram of a single-inductor, bipolar-output boost converter operating in two-cycle mode  501  according to an embodiment of the present invention includes waveforms for signals PWMI  502 , PWMG  503 , switch current ISW  504 , inductor current IL  505 , positive output diode current IDP  506 , negative output diode current IDN  507 , positive DC output voltage +VDC  508  and negative DC output voltage −VDC  509 . The description of the signals and their relationships is given in the two-cycle mode description for  FIG. 4 . The timing diagram  501  shows one possible result and a person skilled in the art will recognize that the relative timing and signal level scales can be adjusted over a large range with no change in fundamental circuit behavior. The inductor  406  of the boost converter  401  could operate in continuous mode as shown by trace  505 . A person skilled in the art will recognize that the inductor  406  of the boost converter  401  can also operate in discontinuous mode as defined in the literature. 
         [0038]    Referring now to  FIG. 6 , a timing diagram of a single-inductor, bipolar-output boost converter operating in single-cycle mode  601  according to an embodiment of the present invention includes waveforms for signals PWMI  602 , PWMG  603 , switch current ISW  604 , inductor current IL  605 , positive output diode current IDP  606 , negative output diode current IDN  607 , positive DC output voltage +VDC  608  and negative DC output voltage −VDC  609 . The description of the signals and their relationships is given in the single-cycle mode description for  FIG. 4 . The timing diagram  601  shows one possible result and a person skilled in the art will recognize that the relative timing and signal level scales can be adjusted over a large range with no change in fundamental circuit behavior. The inductor  406  of the boost converter  401  could operate in continuous mode as shown by trace  605 . A person skilled in the art will recognize that the inductor  406  of the boost converter  401  can also operate in discontinuous mode as defined in the literature. 
         [0039]    Referring now to  FIG. 7 , a tandem single-inductor, bipolar-output boost converter  701  according to a second embodiment of the present invention includes a DC input  702 , a first input switch PWMI 1   703 , a first input switch current ISW 1   730 , a first ground switch PWMG 1   708 , a first negative output diode  704 , a first negative output diode current IDN 1   731 , a negative output DC voltage  705 , a negative output capacitor  707 , an inductor  706 , a first positive output diode  709 , a first positive output diode current  732 , a positive output DC voltage  710 , a positive output capacitor  711 , a second input switch PWMI 2   723 , a second input switch current ISW 2   733 , a second ground switch PWMG 2   728 , a second negative output diode  724 , a second negative output diode current IDN 2   734 , a second positive output diode  729 , a second positive output diode current  735  and a ground terminal  713 . Components  702 ,  703 ,  704 ,  705 ,  706 ,  707 ,  708 ,  709 ,  710  and  711  comprise a first single-inductor, bipolar-output boost converter. Components  723 ,  724 ,  705 ,  726 ,  707 ,  728 ,  729 ,  710  and  711  comprise a second single-inductor, bipolar-output boost converter. Components  702 ,  705 ,  707 ,  710  and  711  are shared between the two converters. The preferred mode of operation is for each converter to operate in single-cycle mode as described for the single-inductor, bipolar-output boost converter. However, the timing of the switches of the second converter is delayed relative to the first by half of the single-cycle PWM period resulting in inductors  706  and  726  being charged and discharged at different times. The benefit of this mode is apparent in the plots in  FIG. 8  for positive output voltage +VDC  705  or  814  and negative output voltage −VDC  710  or  815 . There is a higher ripple frequency at the output, which reduces the size of the capacitor required to filter the ripple voltage present at the two outputs. A significant reduction in component current performance is also achieved. Each transistor switch, diode and inductor of the tandem boost converter  701  now carries one-half the current that would occur in an equivalent non-tandem, single-inductor, bipolar-output boost converter  401 . Thus a greater range of less expensive components may be used in implementing tandem single-inductor, bipolar-output boost converter  701 . Those skilled in the art will realize that the number of tandem boost converter stages shown in  FIG. 7  (two) could be increased as desired to four or eight, or even beyond to a plurality of converter stages. Each would be coupled to the DC input terminal, the negative DC output terminal, and the positive DC output terminal as shown in  FIG. 7 . Alternative input connections to the tandem boost converter  701  can be implemented by a person skilled in the art by connecting switches PWMG 1   708  and PWMG 2   728  to a negative input source and switches PWMI 1   703  and PWMI 2   728  to ground  713  resulting in a single inductor converter utilizing a negative input voltage. The first diode  704 , second diode  709 , third diode  724  and fourth diode  729  could be replaced with switches to implement a synchronous rectification for both positive  710  and negative  705  outputs. Relative timing of the switches and diode currents can be adjusted by one skilled in the art to optimize overall system efficiency without changing the basic operation of the tandem boost converter  701 . 
         [0040]    Referring now to  FIG. 8 , a timing diagram of a single-inductor, bipolar-output boost converter operating in two-cycle mode  501  according to a first embodiment of the present invention includes waveforms for signals PWMI 1   802 , PWMG 1   803 , first switch current ISW 1   804 , first inductor current IL 1   805 , first positive output diode current IDP 1   806 , first negative output diode current IDN 1   807 , signals PWMI 1   808 , PWMG 1   809 , second switch current ISW 1   810 , second inductor current IL 1   811 , second positive output diode current IDP 1   812 , second negative output diode current IDN 1   813 , positive DC output voltage +VDC  814  and negative DC output voltage −VDC  815 . The description of the signals and their relationships is given in the description for  FIG. 7 . The timing diagram  801  shows one possible result and a person skilled in the art will recognize that the relative timing and signal level scales can be adjusted over a large range with no change in fundamental circuit behavior. The first inductor  706  and the second inductor  726  of the boost converter  701  could operate in continuous mode as shown by trace  805 . A person skilled in the art will recognize that the first inductor  706  and second inductor  726  of the tandem boost converter  701  can also operate in discontinuous mode as defined in the literature. 
         [0041]    Referring now to  FIG. 9 , a boost converter inductor current indirect measurement system  901  according to a first embodiment of the present invention includes a DC input  902 , an input switch PWMI  903 , a ground switch PWMG  909 , a negative output diode  905 , a negative output DC voltage  906 , a negative output capacitor  908 , an inductor  907 , a positive output diode  911 , a positive output DC voltage  912 , a positive output capacitor  913 , an inductor current IL  914 , an inductor positive voltage VLP  904 , an inductor negative voltage VLN  910  and a ground terminal  915  all forming a single-inductor, bipolar boost converter  940 , and an ADC  920 , an ADC output  921 , a digital calculation process  922 , a switch control input PWMI  932 , a switch control input PWMG  933 , a calculated inductor input current  930  and a calculated inductor output current  931 . The single-inductor, bipolar boost converter  940  operates in either single-cycle or two-cycle mode as previously described. ADC  920  measures and subtracts the voltages at VLP  904  and VLN  910  and converts the result into a digital code at ADC output  921 . The code represents the voltage across inductor  907  at the time of ADC data conversion. Signal PWMI  932  is active when switch PWMI  903  is closed. Signal PWMG  933  is active when switch PWMG  909  is closed. Digital calculation process  922  measures the time that both PWMI  932  and PWMG  933  are simultaneously active as the charging time of inductor  907 . The calculation process  922  then detects the average voltage measured by ADC  920  during the charging time. Calculation process  922  multiplies the measured inductor voltage by the charging time and divides by the known value of inductor  907 . The result of this calculation is the maximum current IL  914  through inductor  907 . 
         [0042]    Since the current is a sawtooth waveshape, the average current is calculated by dividing the peak inductor current IL  914  by a factor of two and then multiplying by the charging time divided by the total of the charging time and non-charging time for one cycle. The result is the inductor input current  930 . The time that the inductor is not charging during one operating cycle is the non-charging time. The calculation process  922  then detects the average voltage measured by ADC  920  during the non-charging time. Calculation process  922  multiplies the measured inductor voltage by the non-charging time and divides by the known value of inductor  907 . The result of this calculation is the maximum current IL  914  through inductor  907 . Since the current is a sawtooth waveshape, the average current is calculated by dividing the peak inductor current IL  914  by a factor of two and then multiplying by the non-charging time divided by the total of the charging time and non-charging time for one cycle. The result is the inductor output current  931 . There are many methods by which the ADC and digital calculation process may be implemented. A person skilled in the art will recognize that some or all of the processing defined for the ADC  920  and digital calculation process  922  may be implemented via analog signals, sampled analog signals or digital signals and any combination thereof. The ADC output signal  921  can be a parallel digital word a digital bit-stream or a series of sampled analog voltages or currents. 
         [0043]    Referring now to  FIG. 10 , a multi-phase output buck converter  1001 , according to the prior art, includes a positive DC high voltage input  1002 , a negative DC high voltage input  1003 , a first positive switch  1010 , a first negative switch  1011 , a first inductor  1012 , a first capacitor  1014 , an output L 1   1013 , a second positive switch  1020 , a second negative switch  1021 , a second inductor  1022 , a second capacitor  1024 , an output L 2   1023 , a third positive switch  1030 , a third negative switch  1031 , a third inductor  1032 , a third capacitor  1034  and an output L 3   1033 . The positive DC high voltage input  1002 , negative DC high voltage input  1003 , first positive switch  1010 , first negative switch  1011 , first inductor  1012 , first capacitor  1014  and output L 1   1013  comprise a single buck converter according to the prior art. The circuitry associated with the other two outputs L 2   1023  and L 3   1033  are two similar buck converters according to the prior art. The three converters operate with switch timing such that the resulting waveforms on L 1   1013 , L 2   1023  and L 3   1033  are at different phases in relation to each other. Inductors  1012 ,  1022  and  1032  and capacitors  1014 ,  1024  and  1034  are used to perform filtering on the switching waveforms generated by switches  1010 ,  1011 ,  1020 ,  1021 ,  1030  and  1031 . The currents in the inductors will be a combination of a short-term average, or DC, and high frequency effects. 
         [0044]    Referring now to  FIG. 11 , a buck converter inductor current measurement system  1101  according to a first embodiment of the present invention includes a positive DC high voltage input  1102 , a negative DC high voltage input  1105 , a positive switch  1103 , a negative switch  1104 , an inductor  1107 , a capacitor  1109 , an output  1108 , an inductor input voltage  1106 , an ADC  1120 , an ADC output  1121 , a digital low-pass filter  1122  and a measured short-term average inductor current output  1123 . Positive DC high voltage input  1102 , negative DC high voltage input  1105 , positive switch  1103 , negative switch  1104 , inductor  1107 , capacitor  1109 , output  1108 , and inductor input voltage  1106  comprise a single buck converter stage that can be used as part of a multi-phase output buck converter. ADC  1120  continuously measures the voltage across inductor  1107  by subtracting the measured voltages at inductor input  1106  and the output  1108 . The ADC converts the inductor  1107  voltage to an equivalent digital signal at ADC output  1121 . The digital low-pass filter  1122  removes high frequency switching effects resulting in a short-term average at its output  1123 . The output is a representation of the short-term average voltage across the inductor  1107 . The output can be divided by the known DC resistance of the inductor  1107  to calculate current through inductor  1107 . A person skilled in the art will recognize that some or all of the processing defined for the ADC  1120  and digital calculation process  1122  may be implemented via analog signals, sampled analog signals or digital signals and any combination thereof. The ADC output signal  1121  can be a parallel digital word a digital bit-stream or a series of sampled analog voltages or currents. 
         [0045]    Referring now to  FIG. 12 , a digital pulse-width modulation generator  1201 , according to the prior art, includes a pulse width data input  1202 , a clock input  1203  a multi-bit counter  1209 , a counter output  1204 , three data comparators  1205 ,  1212  and  1218 , three data comparator outputs  1206 ,  1213  and  1219 , two RS flip-flops  1207  and  1214 , a data constant of one  1216 , a data constant of one output  1217 , a maximum count constant  1210 , a maximum count output  1211 , a PWM output  1208  and a PWM clock output  1215 . Assuming that the multi-bit counter  1209  and RS flip-flops  1207  and  1214  have been reset, the initial state of the system is that counter output  1204  is at zero, PWM output  1208  is asserted and PWM clock output is asserted. Applying a continuous stream of clock pulses at clock input  1203  results in the counter output to increment by a value of one on each clock pulse. This process continues until data comparator  1205  detects that counter output  1204  is equal to the pulse width data input  1202  and the comparator output  1206  is asserted resulting in a reset of RS flip-flop  1207  and a corresponding de-assertion of PWM output  1208 . Thus the width of the asserted pulse at PWM output  1208  is defined by the value of the pulse width data input multiplied by the period of the signal at clock input  1203 . Counter  1209  continues incrementing its output  1204  until comparator  1212  detects that counter output  1204  is equal to the maximum count value on maximum count constant output  1211  and asserts comparator  1212  output  1213  resulting in a reset of counter  1209  and a set of RS flip-flops  1207  and  1214 . At this point, PWM output  1208  and PWM clock output  1215  are asserted. The counting cycle then repeats. Whenever the counter output  1204  equals the constant one  1216  via constant one output  1217 , comparator  1218  asserts its output  1219  and resets RS flip-flop  1214  resulting in a de-assertion of PWM clock output  1215 . Thus the PWM clock output  1215  is asserted for one input clock cycle. PWM clock output  1215  is used as a handshake signal to request an updated value of pulse width data  1202  from its associated drive circuitry. The total period of both PWM output  1208  and PWM clock output  1215  is equal to the maximum count constant  1210  times the period of the signal at clock input  1203 . The PWM output  1208  pulse width has a minimum assertion time of zero and a maximum assertion time of the PWM output  1208  period. The resolution of the width of the PWM output  1208  is limited to the period of the signal at clock input  1203 . In practical situations, this resolution is insufficient to meet the needs of low-distortion PWM generated waveforms and can only be improved by increasing the clock input  1203  frequency, maximum count constant  1210  value and multi-bit counter  1209  number of bits resulting in an increase in circuit dimensions, power consumption and overall accuracy requirements. There exist many different methods of implementing the digital pulse-width modulator in the prior art that have essentially the same input and output terminal characteristics utilizing digital circuitry. 
         [0046]    Referring now to  FIG. 13 , a delta-sigma pulse-width modulator  1301 , according to an embodiment of the present invention, includes a pulse width coarse data input  1302 , a pulse width fine data input  1307 , a digital summer  1303 , a digital summer output  1304 , a digital delta-sigma modulator  1308 , a digital delta-sigma modulator output  1309 , a digital pulse-width modulator  1305 , a clock input  1310  and a PWM output  1306 . The digital PWM generator  1305  operates as described for the prior art. The digital delta-sigma modulator  1308  may take on many forms and has been fully described in the prior art. Pulse width coarse data  1302  is added in summer  1303  to delta-sigma modulator output  1309  resulting in summer output  1304 . Summer output  1304  is the pulse width data input into digital PWM generator  1305  and defines the resulting pulse width at PWM output  1306 . Pulse width fine data input  1307  is a multi-bit representation of a fine resolution setting for the overall delta-sigma pulse-width modulator  1201 . Pulse width fine data input  1307  drives delta-sigma modulator  1308  causing its output  1309  on each PWM clock  1311  cycle to be either positive one (+1) or negative one (−1). This results in the digital PWM generator data input  1304  to be incremented or decremented by one with a resulting PWM output  1306  pulse width to be incremented or decremented by one input clock  1310  cycle relative to the value at the pulse width coarse data input  1302 . The delta-sigma modulator output  1309  is encoded using pulse density modulation in which the average output is calculated as the number of times in each state multiplied by the associated state value, then added together and divided by the total number of PWM clock  1311  cycles examined. The encoding also has the property of spectral shaping that improves low-frequency resolution via high frequency dithering of the delta-sigma modulator output  1309 , a well-known property of delta-sigma modulation. The result of applying the method of PWM generation shown in this figure is to achieve a significant equivalent improvement in PWM output resolution without utilizing a higher frequency clock input  1310  to the digital PWM generator  1305  or increasing the PWM clock  1311  output frequency. A person skilled in the art will recognize that there are many possible implementations of the delta-sigma modulator  1308  as defined in the literature. The pulse width fine data could also be defined as an analog voltage or current signal whereby the delta-sigma modulator  1308  would be implemented to perform analog-to-digital conversion. The entire process defined by the delta-sigma pulse with modulator  1301  could be implemented using analog signals, sampled analog signals, digital signals or combinations of analog signals, sampled analog signals and digital signals. The pulse width coarse data input  1302  and pulse width fine data input  1307  could be combined to form a single pulse width input. The delta-sigma modulator can also be replaced by other modulator types to implement other spectral shaping and resolution adjustment functions. The summer  1303  may be implemented as an analog adder or could be implemented as part of the delta-sigma modulator  1308 . The signal  1309  may include more than two states depending on the type of delta-sigma modulator  1038  implementation. 
         [0047]    Having described and illustrated the principle of the invention in a preferred embodiment thereof, it is appreciated by those having skill in the art that the invention can be modified in arrangement and detail without departing from such principles. Although a preferred method and circuit has been shown, the exact details of the preferred method and circuit can be changed as desired as required for a particular application. For example, the partitioning of the described integrated circuit can include or exclude functions shown to reduce power consumption and cost. All signaling inside of the various circuit blocks or between the various circuit blocks could be implemented in analog, sampled analog or digital domains. The input boost converter, charge storage and output buck converter could be combined into a single function by circuit reduction techniques. We therefore claim all modifications and variations coming within the spirit and scope of the following claims.

Technology Classification (CPC): 8