Patent Abstract:
The invention described herein is a novel power circuit capable of electrically driving devices whose mechanical action is accomplished by induced strain transducers. The power circuit neither recycles nor dissipates the energy returned from the near lossless transducer, instead redirects power either to another transducer of the same type or to the transducer itself. The invention is based on a balanced capacitive loading method wherein the load itself acts as the energy storage element in the energy balance system. In the preferred mode, the circuit directs power to a symmetric couple where both loads in the couple consist of one or more transductive elements. The invention eliminates the need for a large power-supply bypass capacitor in driving reactive loads thereby reducing the peak power handled by the d.c. power section.

Full Description:
CROSS-REFERENCE TO RELATED APPLICATIONS  
       [0001]    This application claims benefit under 35 U.S.C. 119(e) from U.S. Provisional Application No. 60/192,887 filed on Mar. 29, 2000. 
     
    
     
       FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT  
         [0002]    None.  
         BACKGROUND OF THE INVENTION  
         [0003]    1. Field of the Invention  
           [0004]    The present invention generally relates to a driver circuit for a variety of symmetric load systems. The invention specifically describes a power circuit based on a balanced capacitive loading method wherein the load itself acts as an energy storage element in the energy balance system. In preferred embodiments, the circuit redirects power from a near lossless load to either another load of the same type or to a capacitively equal inactive element. At any given time, a portion of the symmetric load system acts as an actuator and the remaining pure capacitive portion functions similar to a power bypass capacitor.  
           [0005]    2. Related Arts  
           [0006]    A large class of active control devices incorporate small, high-force transductive mechanisms to develop mechanical force. Electrostriction mechanisms develop mechanical force by the interaction of electric fields within the transducer. Magnetostriction mechanisms develop mechanical force by the interaction of magnetic fields within the transducer. Transductive mechanisms are inherently lossless, therefore the energy pumped into the device is returned except for a small portion expended producing mechanical work.  
           [0007]    Various power circuits are known within the art to drive transductive mechanisms. Linear driver circuits are the most common approach. Linear drivers are very inefficient in that return energy from the transductive mechanism is dissipated thermally and thereby no longer available to drive the mechanism. Some improved performance is obtained with class D implementations of the electronics, however, the issue of how to store the transient return energy remains unresolved.  
           [0008]    A more attractive solution to reverse energy flow is a regenerative driver circuit as disclosed in U.S. Pat. No. 6,001,345 issued to Murray et al. on Jan. 4, 2000. However, the invention by Murray suffers two fundamental problems. First, the invention requires a negative impedance inverter that is both quite complex to achieve and never adequately demonstrated in practice. Secondly, the invention requires a large output bypass capacitor. The capacitor value is chosen according to  
             R   Load   C   Filter &gt;&gt;1 /F    
           [0009]    where F is the ripple frequency. The ripple current is in this case impressed by the transients in and out during switching. This leads to a minimal requirement of the output bypass capacitor, where  
           C filter &gt;&gt;C load    
           [0010]    as to achieve a ω 3db  bypass. Consequently, the power bypass capacitor quickly becomes the dominating factor in terms of mass, volume, and performance at larger loads. The result is diminished advantages in terms of efficient power handling and compact implementation of the switching section in the drive topology.  
           [0011]    The deficiencies of Murray et al. are best represented by example. Two symmetric load devices each consisting of two transductive elements are described in FIGS. 8 and 10. Two symmetric load devices each consisting of four transductive elements are described in FIGS. 11 and 12. For discussion purposes, piezo-mechanical motion is induced by piezoelectric transducers with 5.5 μF capacitance operating at 200 volts peak amplitude. The total resultant piezo-capacitance is 22 μF (4 times 5.5) for dually-opposed implementations and 44 μF (8 times 5.5) for quadrature-opposed implementations. Circulating currents from the large piezo-capacitive load has a deleterious effect on a high-voltage power supply because of high ripple causing ‘gain’ ripple induced at the switching amplifier output. The result is decreased stability at higher loads. A bypass capacitor one-hundred times the piezo-capacitance, 2,200 μF and 4,400 μF respectively for the examples above, is required to lower the return ripple of the recirculating current. Such bypass capacitors must operate satisfactorily within the bandwidth of the system, at a minimum several hundred hertz. Such operating conditions provide serious design challenges for both regenerative and conventional driver circuits.  
           [0012]    Conventional power circuits are designed to drive only one side of a transductive system. When applied to a symmetrically coupled transductive system, the lossless nature of the transducers requires nearly all of the input energy returned and either transferred out the system as thermal energy or recovered and redirected. If recovered, the energy is typically recycled with additional input side energy to drive the other symmetric load at the output side of the circuitry. The recovery-recycle methodology as applied to symmetrically coupled systems by conventional circuits produces large peaks in the power supply ripple current. Consequently, such systems are inherently unstable.  
           [0013]    What is required is a power control circuit capable of rapidly redirecting energy between loads in a symmetrically coupled arrangement and specifically a system wherein said loads are transductive elements. The circuit should substantially reduce peak power loading without increasing total power demand. The circuit should eliminate the large bypass capacitor required in the related arts, thereby facilitating a smaller, lighter package. The circuit should eliminate the power supply related stability problems inherent to regenerative and conventional electroncs.  
         SUMMARY OF THE INVENTION  
         [0014]    A first object of the present invention is to provide a small, lightweight electronic driver circuit eliminating the need for large d.c. power bypass capacitors to drive transducer actuated symmetric reactive load systems.  
           [0015]    A second object of the present invention is to provide more volumetrically efficient d.c. power section by effectively removing peak power requirements, leaving only low-level average power to be serviced.  
           [0016]    A third object of the present invention is to provide regenerative efficiency without the need for a large d.c. power bypass capacitor to drive transducer-actuated systems.  
           [0017]    A fourth object of the present invention is to provide increased stability when electrically powering transducer-actuated symmetric reactive load systems at higher charge levels.  
           [0018]    To these ends, the present invention provides a regenerative class D power circuit attached to a symmetrically terminated reactive load system. The power circuit incorporates a new balanced capacitive loading method using the pure reactive portion of the load itself as an energy storage element in the energy balance system. In the present invention either a half-bridge FET or dual half-bridge FET switching topology controls charge-discharge between the two halves of a symmetric reactive load system. The invention can be implemented in the preferred embodiment consisting of a single half bridge or second embodiment consisting of dual half bridges driven 180 degrees out of phase. The topology of the present invention causes energy to be cycled from one side of the symmetric output load to the other side of the symmetric output load. Half-bridge averaging in the invention is externally commanded via a control module. When half-bridge averaging is commanded, an imbalance is caused producing current to flow in one desired direction only. The invention causes the charge to equilibrate between the two symmetric output loads in reference to the new average control module charge. The load on the driver at any given instant is the total output load, while load on the d.c. power supply is only the real power to the load used plus any switching losses. A control module, one example being a PWM, is employed as to institute power flow between symmetric loads as seen on the output side of the circuit. The present invention optimizes the coupling of energy in the L/C circuit comprising the symmetric loads as seen at the output of the circuitry.  
           [0019]    The present invention minimizes power supply conditioning bypass capacitor requirements. Conventional half-bridge power supply circuits require a large bypass capacitor to filter all of the ripple current related to driving the reactive load. FIG. 1 a  shows a conventional half-bridge arrangement wherein the ripple is related to  
           (X Load +Z load )×I Load    
           [0020]    In the present invention, the circuit is required only to filter the ripple current related to the real power dissipated in driving the compound symmetric, reactive or more specifically capacitive, load. In the present invention, the load is a priori symmetrically divided and this fact is used to terminate the circuit uniquely as shown in FIG. 1 b . Thus, the current ripple is only related to  
           X Load ×I Load    
           [0021]    The present invention offers several key advantages over class C, class D and class D regenerative circuitry. The present invention is lighter and smaller with increased efficiency over the related arts. The present invention significantly reduces the high-voltage power supply bypass capacitor representing the largest component in class D and regenerative class D circuitry. The present invention enables larger effective output filter values in a smaller package thereby increasing robustness. Thus, the present invention enables the compact, lightweight implementation for driving high-voltage symmetric output load systems. The present invention effectively enables higher switching voltage into symmetric output reactive load systems thereby retaining the high efficiency of regenerative drivers.  
           [0022]    The present invention is applicable to a wide range of transductive systems including bimorph mechanisms, inchworm devices examples of which are described in U.S. Pat. Nos. 3,902,084, 3,902,085, 4,874,979, and 5,751,090, quadrature MEMS (micro-electromechanical systems) gearing, piezoelectric powered scroll compressors an example described in U.S. Pat. No. 4,950,135, and piezoelectric activated optical communication devices. The advantage of the present invention is that it substantially reduces the instantaneous loads on the high-voltage power supply. This in turn, significantly reduces the power supply mass and volume. In contrast to power switching electronics in the related arts, the present invention is easily miniaturized due to the elimination of large power filter components. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0023]    The invention will now be described in more detail, by way of example only, with reference to the accompanying drawings, in which:  
         [0024]    [0024]FIG. 1 compares a conventional half-bridge switching mechanism with the present invention.  
         [0025]    [0025]FIG. 2 describes charge transfer between two loads in series.  
         [0026]    [0026]FIG. 3 shows a circuit diagram for a half-bridge embodiment of the present invention.  
         [0027]    [0027]FIG. 4 graphically describes the voltage waveforms for a typical symmetric load system.  
         [0028]    [0028]FIG. 5 graphically describes the energy flow waveforms for a typical symmetric load system.  
         [0029]    [0029]FIG. 6 graphically describes the s witch states for a typical symmetric load system.  
         [0030]    [0030]FIG. 7 is a circuit diagram for a dual half-bridge embodiment of the present invention.  
         [0031]    [0031]FIG. 8 describes a bimorph embodiment of the present invention.  
         [0032]    [0032]FIG. 9 shows a preferred embodiment of the bimorph embodiment.  
         [0033]    [0033]FIG. 10 shows an example inch-worm implementation of the present invention.  
         [0034]    [0034]FIG. 11 shows a generic quadrature implementation of the present invention.  
         [0035]    [0035]FIG. 12 shows an example piezo-electric activated optical communications device incorporating the present invention. 
     
    
     REFERENCE NUMERALS  
       [0036]    SW 1  First switch  
         [0037]    SW 2  Second switch  
         [0038]    L 1  Filter inductor  
         [0039]    C 1  Bypass capacitor  
         [0040]    [0040] 1  Switchmode power control circuit  
         [0041]    [0041] 2  First load  
         [0042]    [0042] 3  Second load  
         [0043]    [0043] 4  Center tap  
         [0044]    [0044] 5  Node B  
         [0045]    [0045] 6  Node C  
         [0046]    [0046] 7  Node D  
         [0047]    [0047] 8  Node E  
         [0048]    [0048] 9  Node F  
         [0049]    [0049] 10  Switch controller circuit  
         [0050]    [0050] 11  Power supply  
         [0051]    [0051] 12  Node H  
         [0052]    [0052] 13  Node I  
         [0053]    [0053] 14  Regenerative drive  
         [0054]    [0054] 16  Ground  
         [0055]    [0055] 17  First single load  
         [0056]    [0056] 18  Second single load  
         [0057]    [0057] 19  Driver circuit  
         [0058]    [0058] 20  Median axis  
         [0059]    [0059] 22  First outer layer  
         [0060]    [0060] 23  Second outer layer  
         [0061]    [0061] 24  First transductive element  
         [0062]    [0062] 25  Second transductive element  
         [0063]    [0063] 26  Middle layer  
         [0064]    [0064] 27  Adhesive  
         [0065]    [0065] 28  Bimorph actuator  
         [0066]    [0066] 30  Planar configuration  
         [0067]    [0067] 31  Upper layer  
         [0068]    [0068] 32  Lower layer  
         [0069]    [0069] 40  Phase differential controller  
         [0070]    [0070] 41  First member  
         [0071]    [0071] 42  Second member  
         [0072]    [0072] 43  First actuator  
         [0073]    [0073] 44  Second actuator  
         [0074]    [0074] 45  Third actuator  
         [0075]    [0075] 46  Fourth actuator  
         [0076]    [0076] 48  Symmetric reactive load system  
         [0077]    [0077] 50  Flex tensioner  
         [0078]    [0078] 51  First tensioner  
         [0079]    [0079] 52  Second tensioner  
         [0080]    [0080] 53  Third tensioner  
         [0081]    [0081] 54  Fourth tensioner  
         [0082]    [0082] 55  First transverse element  
         [0083]    [0083] 56  Center transverse element  
         [0084]    [0084] 57  Second transverse element  
         [0085]    [0085] 58  Mechanical connector  
         [0086]    [0086] 59  Mechanical ground  
       DESCRIPTION OF THE INVENTION  
       [0087]    [0087]FIG. 2 generally describes the present invention at a functional level. The invention consists of a first load  2 , a second load  3 , and a driver circuit  19 . Both first load  2  and second load  3  have identical mechanical and electrical impedance. The driver circuit  19  provides d.c. voltage to both first load  2  and second load  3  arranged in series such that at equilibrium one-half of the total voltage (V) from the power supply  11  within the driver circuit  19  resides within the first load  2  (V/2) and the second load  3  (V/2). This condition is referred to as the equilibrated charge state and is represented in FIG. 2 a.    
         [0088]    The driver circuit  19  cycles and recycles power between the first load  2  and the second load  3  via the charge transfer process. In FIG. 2 b , the driver circuit  19  directs power from the first load  2  to the second load  3 . While the total voltage (V) across first load  2  and second  3  equals the power supply  11  voltage, more voltage resides within the second load  3 . In FIG. 2 c , the driver circuit  19  redirects power from the second load  3  to the first load  2 . Again while the total voltage (V) across first load  2  and second load  3  is equal to the power supply  11  voltage, more voltage now resides within the first load  2 . In both charge flow descriptions, the driver circuit  19  alters current flow within a half-bridge topology via opening (OFF condition) and closing (ON condition) of two switches. During charge transfer, the load from which charge is directed is functionally an energy storage element facilitating the transfer process.  
         [0089]    The switchmode power control circuit  1  including a first load  2  and second load  3  connected to a driver circuit  19 . The driver circuit  19  consists of a filter inductor L 1 , a regenerative drive  14 , a bypass capacitor C 1 , and a power supply  11 . FIG. 3 describes the half-bridge embodiment of the switchmode power control circuit  1 . First load  2  and second load  3  are connected in series at the center tap  4 , node B  5 , and node D  7 . The negative terminal from the first load  2  is connected to the positive terminal from the second load  3  at the center tap  4 . The positive terminal from the first load  2  is connected to node B  5  thereby aligning the positive terminal with the positive output on the power supply  11 . The negative terminal from the second load  3  is connected to node D  7  thereby aligning the negative terminal with the negative output on the power supply  11 . A filter inductor L 1  is connected to the center tap  4  between first load  2  and second load  3  and node C  6  between first switch SW 1  and second switch SW 2 . Nodes B  5 , C  6 , and D  7  facilitate connection of first load  2 , second load  3 , and filter inductor L 1  to the regenerative drive  14 . The regenerative drive  14  consists of a first switch SW 1 , a second switch SW 2 , and a switch controller  10 . First switch SW 1  and second switch SW 2  are connected in series to node B  5 , node D  7 , and dually to node C  6  thereby parallel to both first load  2  and second load  3 . A switch controller circuit  10  is connected to both first switch SW 1  and second switch SW 2 . Parallel to both first switch SW 1  and second switch SW 2  and opposite of both first load  2  and second load  3  is a bypass capacitor C 1  connected at node E  8  and node F  9 . A power supply  11  is connected adjacent to the bypass capacitor C 1 . The power supply  11  is of finite impedance and applies d.c. voltage to the driver circuit  19 . The power supply  11  replenishes voltage lost during switching and that portion expended by first load  2  and second load  3 .  
         [0090]    The regenerative circuit  14  is known within the art. A typical embodiment consists of a first switch SW 1  , second switch SW 2 , and switch controller circuit  10 . Both first switch SW 1  and second switch SW 2  rapidly and alternately switch between OFF and ON, thereby adjusting current flow within the switchmode power control circuit  1  and energy flow between the first load  2  and the second load  3 . Example switches SW 1 , SW 2  include bipolar transistors, MOSFET&#39;s and IBGT&#39;s, all known within the art. The filter inductor L 1  stores energy when either first switch SW 1  or second switch SW 2  is ON thereby providing a temporary charge flow bias at the onset of the next switching condition. The switch controller circuit  10  consists of a high-frequency PWM modulator and driver circuitry known within the art. The switch controller circuit  10  controls timing and duration of OFF and ON conditions at first switch SW 1  and second switch SW 2 . In preferred embodiments, OFF and ON switching at both first switch SW 1  and second switch SW 2  occurs at frequencies in the hundreds of kilohertz. The PWM is modulated with the desired waveform, examples including but not limited to sine, square, and sawtooth waves. The bypass capacitor C 1  compensates for alternating current conditions at the power supply  11  thereby eliminating current ripple. The switchmode power control circuit  1  is terminated to a ground  16  by methods known within the art.  
         [0091]    First load  2  and second load  3  may consist of one or more capacitive elements. In the most preferred embodiment, both first load  2  and second load  3  are mechanically and electrically matched transductive element. While various embodiments are possible, the total mechanical and electrical impedance of the first load  2  closely approximate that of the second load  3 . A second embodiment for non-symmetric loading conditions consists of an approximately chosen passive capacitor whose value added to none or partial existing second load  3  now matches the first load  2  in non-resonant applications.  
         [0092]    [0092]FIG. 4 describes typical voltage waveforms at both first load  2  and second load  3 . FIG. 5 describes typical energy flow waveforms for first load  2  and second load  3 . Both Figures assume a sinusoidal command function from the switch controller circuit  10  into the first switch SW 1  and the second switch SW 2 . However, any fixed or variable function is applicable to the present invention. FIG. 6 artistically describes OFF and ON conditions at first switch SW 1  and second switch SW 2  for waveforms profiles in FIGS. 4 and 5 to aid functional visualization.  
         [0093]    The equilibrated charge state is identified in FIG. 4 as a horizontal line with a magnitude V/2 representing one-half the total voltage (V) across the power supply  11 . This condition is maintained by the rapid OFF and ON switching of first switch SW 1  and second switch SW 2  at a constant frequency of equal duty cycle duration. Neither charge nor discharge occur at the equilibrated charge state. Voltage at center tap  4  is one-half of the power supply  11  voltage (V) and at node C  6  is either the power supply  11  voltage (V) or zero.  
         [0094]    Charge transfer from the second load  3  to the first load  2  is achieved by increasing the duration of the ON condition at the second switch SW 2  thereby causing a corresponding increase in the OFF condition at the first switch SW 1 . Switching bias increases the discharge of energy at the second load  3  facilitating redirection to the first load  2 . Alternatively, charge transfer from the first load  2  to the second load  3  is achieved by increasing the duration of the ON condition at the first switch SW 1  thereby causing a corresponding increase in the OFF condition at the second switch SW 2 . Here, biased switching effectively increases the discharge of energy at the first load  2  and redirects it into the second load  3 . The resultant voltage waveforms for both first load  2  and second load  3  are sinusoidal however phase shifted 180 degrees. The total sum voltage at any time is equal to the power supply  11  voltage (V). The energy flow waveforms for first load  2  and second load  3  are also sinusoidal and phase shifted 180 degrees. Additionally, current and energy flow waveforms for each of the first load  2  and second load  3  are phase shifted 90 degrees.  
         [0095]    The charge transfer process at the circuit level is the following. When the first switch SW 1  is ON and the second switch SW 2  is OFF, current in the filter inductor L 1 , accumulated when the second switch SW 2  was ON charging node C  6  to V and center tap  4  to V/2, continues to flow in the positive direction for a short duration into the first switch SW 1 . Thereafter, the charge direction reverses into the filter inductor L 1  since voltage at node C  6  is now zero and the voltage at center tap  4  is V/2. This charge flow pattern effectively “pulls” current from center tap  4  through first load  2  and second load  3  and “pushes” current into the ground  16 . When the first switch SW 1  is OFF and the second switch SW 2  is ON, current in the filter inductor L 1 , accumulated when the first switch SW 1  was ON causing node C  6  to have no voltage and placing center tap  4  at V/2, continues to flow in the negative direction for a short duration into the second switch SW 2 . Thereafter, the charge direction reverses away from the filter inductor L 1  since voltage at node C  6  is now the power supply  11  voltage (V) and the voltage at center tap  4  is one-half the power supply  11  value. This charge flow pattern effectively “pulls” current from node C  6  and “pushes” current through the first load  2  and second load  3 . But because the loads are not referenced to the same point, the current causes a differential variation in the loads thereby effectively producing the “pushing” and “pulling” described above.  
         [0096]    [0096]FIG. 7 illustrates a dual half-bridge embodiment of the present invention wherein a first single load  17  and a second single load  18  are respectively connected to a filter inductor L 1 , a regenerative drive  14 , and a bypass capacitor C 1  and thereafter connected to a common power supply  11  at node H  12  and node  113 . In this embodiment, charge-discharge at the first single load  17  and second single load  18  are independent of one another. When the first switch SW 1   a  is ON and the second switch SW 2   a  is OFF, charge flows from positive to negative across the first single load  17 . When the first switch SW 1   a  is OFF and the second switch SW 2   a  is ON, charge flows from negative to positive across the first single load  17 . When the first switch SW 1   b  is ON and the second switch SW 2   b  is OFF, charge flows from positive to negative across the second single load  18 . When the first switch SW 1   b  is OFF and the second switch SW 2   b  is ON, charge flows from negative to positive across the second single load  18 . Alternating functionality between first single load  17  and second single load  18  is achieved by phase shifting command functions from the switch controller circuits  10   a  and  10   b . A 180 degree phase shift is implemented in the preferred embodiment. While the single half-bridge shown in FIG. 3 was applicable to capacitive elements, the dual half-bridge embodiment is applicable to either dually arranged inductive or dually arranged capacitive elements.  
         [0097]    Examples 1 through 4 demonstrate implementations of the present invention to symmetric reactive load systems  48 .  
       EXAMPLE I  
       [0098]    [0098]FIG. 8 describes the extension of the present invention to a new device within the class of referred to as bimorphs. The switchmode power control circuit  1  is a mechanical half-bridge in this system. A bimorph actuator  28  consists of a plurality of planar members about a median axis  20 . The preferred embodiment consists of a middle layer  26  sandwiched between a first transductive element  24  and a second transductive element  25 . The middle layer  26  is a material sufficient to isolate the first transductive element  24  from the second transductive element  25 . Transductive elements  24 ,  25  may consist of one or more capacitive elements, however the total capacitance of both transductive elements  24 ,  25  are approximately equal. In the preferred embodiment, a first outer layer  22  and a second outer layer  23  further sandwich the transductive elements  24 ,  25 . The outer layers  22 ,  23  are any stiff yet flexible homogeneous or composite material with the preferred embodiment being a metal. In the most preferred embodiment, the transductive elements  24 ,  25  are bonded to the middle layer  26  and outer layers  22 ,  23 .  
         [0099]    The bimorph actuator  28  forms a planar configuration  30  either when no charge is applied to the transductive elements  24 ,  25  or when equal charges are applied within the switchmode power control circuit  1  to the transductive elements  24 ,  25 , as shown in FIG. 8 a . The planar configuration  30  is altered via the driver circuit  19  by the charge transfer method. Charge transfer is achieved when the charge balance is altered between transductive elements  24 ,  25  resulting in biased displacement of the bimorph actuator  28 , sometimes referred to as the unimorph effect. FIG. 8 b  shows upward curvature in the bimorph actuator  28  about the median axis  20  when charge is removed from the first transductive element  24  and applied to the second transductive element  25 . FIG. 8 c  shows downward displacement in the bimorph actuator  28  about the median axis  20  when charge is removed from the second transductive element  25  and applied to the first transductive element  24 . Charge flow directions are noted in FIGS. 8 b  and  8   c . The amount of displacement is limited by the charge saturation characteristics of the transductive elements  24 ,  25  and the stiffness of the bimorph actuator  28 .  
         [0100]    [0100]FIG. 9 shows a preferred embodiment of the bimorph actuator  28  functioning as an actuator. The pre-stressed bimorph actuator  28  consists of a steel or titanium middle layer  26 , a piezoceramic first transductive element  24 , a piezoceramic second transductive element  25 , an aluminum first outer layer  22 , and an aluminum second outer layer  23  wherein layers  22 ,  23 ,  26  and elements  24 ,  25  are bonded by an adhesive  27 . In other embodiments, an upper layer  31  and a lower layer  32  are applied to the bimorph actuator  28  consisting of a low-friction material preferably polytetrafluoroethylene. The most preferred embodiment consisting of the following: outer layers  22 ,  23  being a 1.96 inch wide by a 1.96 inch long by a 0.001 inch thick aluminum, ASTM B20, plate; transductive elements  24 ,  25  being a 2.04 inch wide by 2.04 inch long by 0.015 inch thick 3195HD ceramic manufactured by the CTS Corporation of Albuquerque, N. Mex.; middle layer  26  being a 3.0 inch wide by 2.24 inch long by 0.02 inch thick stainless steel plate, type 302, ASTM A117; and adhesive  27  being a high temperature polyimide commonly known as LaRC-SI.  
         [0101]    The preferred embodiment is assembled with the following process. The outer layers  22 ,  23  are perforated and cleaned. The piezoceramics are cleaned and sprayed with LaRC-SI solution (e.g., 8% LaRC-SI powder and 92% N-methyl-pyrolidinone) and then dried in an oven. The middle layer  26  is scuffed, primed, piezoceramics applied to the middle layer  26 , and outer layers  22 ,  23  applied to the piezoceramics. A pre-heat step may be used to soften the adhesive  27  and provide the adherence required to keep elements  24 ,  25  and layers  22 ,  23 ,  26  together during assembly. An alcohol solution also serves the same purpose. To insure a uniform bond, a vacuum bagging process is used to plate and fixture as to apply equal pressure onto individual elements while in the autoclave. The bimorph actuator  28  is placed into the autoclave, platen pressed, and subject to a pressure and temperature. During the autoclave cycle, the bimorph actuator  28  is heated, squeezed, cooked, then cooled to room temperature. During cool down, differences in the thermal coefficients of expansion between metals and ceramic creates a stress state within the material resulting in a flat planar configuration  30 .  
         [0102]    The bimorph actuator  28  is polarized on either the outside of each ceramic or on the top of each ceramic. Three wires are attached to the structure. One wire is attached to the first outer layer  22  thereby providing a positive. A second wire is attached to the second outer layer  23  thereby providing a negative. And a third wire is attached to the middle layer  26  for grounding.  
         [0103]    A multilaminar version of the bimorph actuator  28  is realized by the sequential layering of two or more bimorph actuators  28  separated by a frictionless material as described by the upper layer  31  and the lower layer  32 . Two electroding options are possible. The first option consists of similarly poling and driving the piezoceramics in parallel on one side of the median axis  20 , thereby functioning as the first load  2 , and similarly poling and driving the piezocermics in parallel on the opposite side of the median axis  20 , thereby functioning as the second load  3 . The second option alternates poling and electroding thereby treating odd numbered piezoceramics as the first load  2  and even numbered piezoceramics as the second load  3 .  
       EXAMPLE II  
       [0104]    [0104]FIG. 10 describes the application of the present invention to a symmetric reactive load system  48  consisting of two identical but opposed induced strain devices. Such systems are referred to as inchworm or bi-static motion devices, and generally identified as two-state machines. The driver circuit  19  functions as a symmetric mechanical half-bridge between a first transductive element  24  and a second transductive element  25 . The transductive elements  24 ,  25  are typically induced strain transducers having equal mechanical and electrical impedance.  
         [0105]    [0105]FIG. 10 a  shows a typical two-state machine either when no charge is applied to the transductive elements  24 ,  25  or when equal charge is applied to the transductive elements  24 ,  25  via the driver circuit  19 . This condition is referred to as the equilibrated charge state. In this example, the length of the transductive elements  24 ,  25  are altered by redirecting charge, thereby altering the equilibrated charge state, between the transductive elements  24 ,  25  via the driver circuit  19 . In FIG. 10 b , charge is removed from the first transductive element  24  and directed into the second transductive element  25 . As charge flows from the first transductive element  24  there is a decrease in the induced mechanical strain within the transductive element  24  thereby causing mechanically contractive displacement. As charge flows into the second transductive element  25  there is an increase in the induced mechanical strain within the transductive element  25  thereby causing a mechanically extensive displacement. The degree of displacement is limited by the charge saturation limit of the second transductive element  25 . To reverse the mechanical effect, the driver circuit  19  removes charge from the second transductive element  25  and returns it to the first transductive element  24 , as shown in FIG. 10 c . As the charge on the second transductive element  25  decreases, the mechanical strain causing its extension is reduced causing a corresponding decrease in its length. As the charge on the first transductive element  24  increases, mechanical strain within the transductive element  24  increases causing its extension. The degree of displacement is limited by the charge saturation limit of the first transductive element  24 . The driver circuit  19  and its application herein operates quasi-statically, over varying frequency or at resonance and applicable to piezo-motors, precision positioners, z-y stepping systems, and piezo-based differential control isolation systems.  
       EXAMPLE III  
       [0106]    [0106]FIG. 11 describes the application of the present invention to a symmetric reactive load system  48  consisting of a quadrature arrangement of equal reactance transductive mechanisms sharing identical mechanical and electrical impedance. A first actuator  43 , a second actuator  44 , a third actuator  45 , and a fourth actuator  46  are arranged in a quadrature to produce motion between a first member  41  and a second member  42  via the linear extension and contraction of the actuators  43 ,  44 ,  45 ,  46 . Orbital motion between first member  41  and second member  42  is applicable to scroll compressors. Rotational motion between the first member  41  and second member  42  is applicable to MEMS gearing. The actuators  43 ,  44 ,  45 ,  46  are typically induced strain transducers. One embodiment of the symmetric reactive load system  48  consists of two orthogonal actuators, for example the first actuator  43  and the second actuator  44 . Another embodiment consists of two pairs of orthogonal arranged actuators  43 ,  44 ,  45 ,  46 , as shown in FIG. 11. In the latter embodiment, the first actuator  43  and third actuator  45  are driven in parallel, as well as the second actuator  44  and fourth actuator  46 .  
         [0107]    The extension and contraction of the actuators  43 ,  44 ,  45 ,  46  are achieved via the driver circuit  19  using the charge transfer method described in Example II. The equilibrated charge state exists either when the charge to the first actuator  43  and the third actuator  45  are equal and the charge to the second actuator  44  and the fourth actuator  46  are equal or when no charge resides in all four actuators  43 ,  44 ,  45 ,  46 . A phase differential controller  40  insures a ninety-degree phase shift at the third actuator  45  and the fourth actuator  46 .  
         [0108]    The driver circuit  19  avoids driving actuator pairs  43  and  45 ,  44  and  46  at piezo-resonance via a dual half-bridge topology, thereby developing only mechanical resonance. This mechanical resonance approach uses a single half-bridge output stage in a self-oscillatory system avoiding direct coupling between the energy in the resonant circuit and pressure in the system. The resultant system either operates in an electrically resonant mode or a electrical-mechanical resonant mode. The direct drive mode possesses a simpler method of feedback control than the non-resonant mode. It should be noted that in the direct drive mode the actuators  43 ,  44 ,  45 ,  46  must operate in a bipolar voltage mode.  
       EXAMPLE IV  
       [0109]    [0109]FIG. 12 describes the application of the present invention to a two-state bidirectional device referred to as a flex tensioner  50 . A typical flex tensioner  50  consists of four identical but dually opposed induced strain mechanisms referred to as a first tensioner  51 , a second tensioner  52 , a third tensioner  53 , and a fourth tensioner  54 . Each tensioner  51 , 52 ,  53 ,  54  may consist of one or more transductive elements  24 ,  25 . Extension and contraction of the tensioners  51 ,  52 ,  53 ,  54  are achieved and coordinated via the driver circuit  19  using the charge transfer method described in Example II. An equilibrated charge state exists either when equal charge is applied to each of the tensioners  51 ,  52 ,  53 ,  54  or when no charge is present in all tensioners  51 ,  52 ,  53 ,  54 .  
         [0110]    In the method of operation of the present invention, the driver circuit  19  transfers charge from first tensioner  51  and fourth tensioner  54  to the second tensioner  52  and third tensioner  53 . During charge transfer, an external charge is applied via the driver circuit  19  to maintain the sum total charge in the tensioners  51 ,  52 ,  53 ,  54 . Charge depletion in the first tensioner  51  and the fourth tensioner  54  causes the first transverse element  55  and second traverse element  57  to expand in a mechanically amplified fashion due to outward displacement. Simultaneously, charge addition in the second tensioner  52  and the third tension  53  causes the center transverse element  56  to compress along its length in a mechanically amplified fashion due to inward displacement. Mechanical connectors  58  between first tensioner  51  and second tensioner  52  and between third tensioner  53  and fourth tensioner  54  insure compression of the center transverse element  56 . Mechanical grounds  59  are positioned along the flex tensioner  50 .  
         [0111]    Extension of the center transverse element  56  is achieved by diverting charge flow from second tensioner  52  and third tensioner  53  to first tensioner  51  and fourth tensioner  54  via the driver circuit  19 . First tensioner  51  and fourth tensioner  54  produce inward displacement adjacent to the first transverse element  55  and the second transverse element  57 . Second tensioner  52  and third tensioner  53  produce outward displacement adjacent to the center transverse element  56 . The magnitude of compression and extension within the center transverse element  56  is voltage dependent. Voltage and current are monitored at each tensioner  51 ,  52 ,  53 ,  54 . Values are analyzed by the driver circuit  19  so that voltage levels are maintained and so that charge compensation is provided to offset mechanical and switching losses.  
         [0112]    The description above indicates that a great degree of flexibility is offered in terms of the present invention. Although the present invention has been described in considerable detail with reference to certain preferred versions thereof, other versions are possible. Therefore, the spirit and scope of the appended claims should not be limited to the description of the preferred versions contained herein.

Technology Classification (CPC): 8