Patent Abstract:
Circuits and methods relating to the provision of a reactive current to ensure zero voltage switching in a boost power factor correction converter. A simple passive circuit using a series connected inductor and capacitor are coupled between two phases of an interleaved boost PFC converter. The passive circuit takes advantage of the 180° phase-shift between the two phases to provide reactive current for zero voltage switching. A control system for adjusting and controlling the reactive current to ensure ZVS for different loads and line voltages is also provided.

Full Description:
TECHNICAL FIELD 
     The present invention relates to circuits and power supplies. More specifically, the present invention relates to circuits and methods for use in AC/DC converters which can be used to ensure zero voltage switching. 
     BACKGROUND OF THE INVENTION 
     The ever growing popularity of environmentally friendly or “green” products and services has led to a renewed interest in the electric car. While the practical, fully electric powered car might still be a few years in the future, its cousin, the hybrid car is already being used by the general public. Both these types of vehicles have their batteries usually recharged by plugging into regular 120V or 220V wall sockets. Developments that affect battery charging systems would therefore have a great impact on these types of vehicles. 
     Electric Vehicle (EV) power conditioning systems usually utilize a high energy battery pack to store energy for the electric traction system. A typical block diagram of the power conditioning system in an EV is shown in  FIG. 1 . The high energy battery pack is charged from a utility AC outlet. This energy conversion during the battery charging is performed by an AC/DC converter. Such AC/DC converters include a front-end boost converter, which performs input power factor correction (PFC) and AC/DC conversion, and a full-bridge DC/DC converter, for battery charging and galvanic isolation. PFC is useful for improving the quality of the input current, which is drawn from the power utility so as to comply with the regulatory standards like IEC1000-3-2. 
     Switching losses of the power switches in boost PFC AC/DC converters significantly deteriorate the efficiency of the converter. Present products usually use active auxiliary circuits in order to provide soft-switching, which increases the complexity of the system while decreasing its reliability. 
     One technique which can reduce switching losses is zero voltage switching. Most converters use MOSFETs (metal oxide semiconductor field effect transistors) in low to medium power applications (i.e. application involving a few kilowatts). In order to have robust and reliable operation, MOSFETs are preferably switched under zero voltage. Operating at Zero Voltage Switching (ZVS) decreases the converter switching losses and provides a noise free environment for the system control circuit. Loss of ZVS means extremely high switching losses at high switching frequencies and very high EMI. Loss of ZVS can also cause a very noisy control circuit, which leads to shoot-through and loss of the semiconductor switches. 
     There is therefore a need for solutions which lessen switching losses without the drawbacks of the prior art. 
     SUMMARY OF INVENTION 
     The present invention provides circuits and methods relating to the provision of a reactive current to ensure zero voltage switching in a boost power factor correction converter. A simple passive circuit using a series connected inductor and capacitor are coupled between two phases of an interleaved boost PFC converter. The passive circuit takes advantage of the 180° phase-shift between the two phases to provide reactive current for zero voltage switching. A control system for adjusting and controlling the reactive current to ensure ZVS for different loads and line voltages is also provided. 
     In a first aspect, the present invention provides a circuit for use in a power factor correction (PFC) converter circuit having a full bridge diode rectifier subcircuit, the circuit comprising:
         first inductor coupled between a positive input node and a first intermediate node;   a second inductor coupled between said positive input node and a second intermediate node;   a first power transistor having a drain lead coupled to said first intermediate node and having a source lead coupled to a negative power node;   a second power transistor having a source lead coupled to said first intermediate node and a drain lead coupled to a positive output node;   a third power transistor having a drain lead coupled to said second intermediate node and having a source lead coupled to said negative power node;   a fourth power transistor having a source lead coupled to said second intermediate node and a drain lead coupled to said positive output node; and   an auxiliary inductor and an auxiliary capacitor coupled in series to one another between said first intermediate node and said second intermediate node;       

     wherein
         said full bridge rectifier subcircuit is coupled between said positive input node and said negative power node;   a load to said circuit is coupled between said positive output node and said negative power node.       

     In a second aspect, the present invention provides a power factor correction AC/DC converter circuit comprising:
         a full bridge diode rectifier subcircuit for receiving an input AC signal;   a first boost converter subcircuit;   a second boost converter subcircuit; and   a passive auxiliary subcircuit coupled between said first and second boost subcircuits, said passive auxiliary subcircuit being for providing reactive current to said boost converter subcircuits for zero voltage switching;       

     wherein
         said first and second boost converter subcircuits are coupled in parallel to said rectifier subcircuit;   said passive auxiliary subcircuit comprises an auxiliary inductor and an auxiliary capacitor coupled in series to each other.       

    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The embodiments of the present invention will now be described by reference to the following figures, in which identical reference numerals in different figures indicate identical elements and in which: 
         FIG. 1  is a block diagram of a typical power conditioning system for electric vehicles; 
         FIG. 2  is a circuit diagram of a circuit according to one aspect of the invention; 
         FIG. 2A  is a block diagram of the circuit of  FIG. 2 ; 
         FIG. 3  illustrates timing waveforms for the circuit in  FIG. 2  for duty cycles greater than 50%; 
         FIG. 4  illustrates timing waveforms for the circuit in  FIG. 2  for duty cycles less than 50%; 
         FIG. 5  shows current waveforms for the auxiliary circuit used in the circuit of  FIG. 2 ; 
         FIG. 6  is a block diagram of a control system for the circuit in  FIG. 2 ; 
         FIG. 7  is a system block diagram of an implementation of one aspect of the invention; 
         FIG. 8  illustrate the current and voltage waveforms for the circuit in  FIG. 2 ;  FIG. 8  and  FIG. 9  show the waveforms of the ZVS boost PFC converter 
         FIG. 9  is an enlargement of  FIG. 8  and shows zero voltage switching; 
         FIG. 10  illustrates waveforms for the two phases of the circuit in  FIG. 2  along with the auxiliary circuit current; 
         FIG. 11  shows the auxiliary inductor current and how it changes with changes in the input current; 
         FIG. 12  illustrates the input voltage and the input current for full-load for the circuit illustrated in  FIG. 2 ; 
         FIG. 13  illustrates efficiency curves for a conventional interleaved boost PFC converter as well as efficiency curves for the circuit in  FIG. 2 . 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Referring to  FIG. 1 , a block diagram of a typical power conditioning system in an electric vehicle is illustrated. As can be seen, a wall plug is coupled to a PFC converter block which, in turn, is coupled to a DC/DC converter. The -PFC converter block and the DC/DC converter block, taken together, form the AC/DC converter block. The output of the AC/DC converter block is then received by a DC/DC converter and a DC/AC converter which, in turn, powers a 3 phase motor. 
     One aspect of the invention may be used in the PFC converter illustrated in  FIG. 1 . This aspect of the invention provides for a ZVS interleaved boost PFC AC/DC converter as described below. This aspect of the invention may be used in substantially any AC/DC converter application. It is, however, particularly useful in Electric Vehicle (EV) applications due to its superior performance and high efficiency. There are two main sources of losses in boost PFC converters: turn-on losses of the boost power switch and reverse recovery losses of the output diode. These losses deteriorate the efficiency and performance of the converter. This aspect of the invention provides soft switching for power semiconductors through an auxiliary circuit placed between the two phases of the converter. The auxiliary circuit provides reactive current to charge and discharge the output capacitors of the power semiconductors to achieve soft-switching and thereby minimize switching losses of the converter. 
     One aspect of the invention also maintains ZVS for the universal input voltage (85 Vrms to 265 Vrms), which includes a very wide range of duty ratios (0.07-1). In addition, the control system aspect of the invention optimizes the amount of reactive current required to guarantee ZVS during the line cycle for different load conditions. This optimization is useful as the converter may work at very light loads for a long period of time. 
       FIG. 2  illustrates a power circuit of the ZVS interleaved boost PFC converter according to one aspect of the invention. The circuit  10  has an input signal from an AC current source  20  (such as a wall plug). This signal is received by a full bridge diode rectifier  30  which is coupled between a positive input node  40  and a negative power node  50 . Coupled between the positive input node  40  and a first intermediate node  60  is a first inductor  70 . Coupled between the first intermediate node  60  and the negative power node  50  is a first power transistor  80 . In a MOSFET implementation of the power transistor, the drain lead is coupled to the first intermediate node  60  and the source lead is coupled to the negative power node  50 . A second power transistor  90  is coupled between the first intermediate node  60  and a positive output node  100 . The second power transistor, in a MOSFET implementation, has its drain lead coupled to the positive output node  100  and its source lead coupled to the first intermediate node  60 . 
     The circuit  10  in  FIG. 2  also has a second inductor  110  coupled between the positive input node  40  and a second intermediate node  120 . A third power transistor  130  is coupled between the second intermediate node  120  and negative power node  50 . For a MOSFET implementation of the third power transistor, its source lead is coupled to the negative power node  50  while its drain lead is coupled to the second intermediate node  120 . A fourth power transistor  140  is coupled between the positive output node  100  and the second intermediate node  120 . For the MOSFET implementation, the fourth power transistor has its drain lead coupled to the positive output node  100  and its source lead is coupled to the second intermediate node  120 . As can be seen, each power transistor (or boost power transistor) has a snubber capacitor and a diode associated with it. For this circuit, the output capacitors, are the combination of each MOSFETs&#39; output capacitors and the snubber capacitors in parallel with MOSFETs, placed to remove turn-off losses. 
     The circuit  10  further has an auxiliary inductor  150  and an auxiliary capacitor  160  coupled in series to one another. This series connected auxiliary circuit is coupled between the first intermediate node  60  and the second intermediate node  120 . The load  170  to the circuit  10  is coupled between the positive output node  100  and the negative power node  50 . 
     Referring to  FIG. 2A , the circuit  10  can be seen as having two boost converter subcircuits  180 A,  180 B coupled to a full bridge diode rectifier subcircuit  30 . The auxiliary subcircuit  200  is coupled between the two boost converter subcircuits  180 A,  180 B. Of course, the auxiliary subcircuit  200  has the auxiliary inductor  150  and auxiliary capacitor  160 . 
     In this circuit, two boost converters subcircuits operate with a 180° phase-shift in order to reduce the input current ripple. This 180° phase-shift can be used to provide reactive current in order to implement ZVS for the power MOSFETs. A simple passive auxiliary subcircuit  200  is used to provide reactive current as shown in  FIGS. 2 and 2A . This auxiliary subcircuit  200  consists of a high frequency inductor  150  and a DC-blocking capacitor  160 . The DC-blocking capacitor is used to avoid having DC current flow through the auxiliary subcircuit  200 . Since there may be a slight difference between the duty ratios of the two phases, this capacitor is necessary to eliminate any DC current arising from the mismatch of the duty ratios of the main switches in the practical circuit. 
       FIG. 3  shows the waveforms for the converter circuit for duty cycles greater than 50% while  FIG. 4  shows the waveforms for the converter for duty cycles less than 50%. The characteristics and benefits of the circuit can be seen from these waveforms. According to these figures, there are eight operating modes in one switching cycle of the converter. 
     From  FIGS. 3 and 4 , it can be seen that the circuit provides a very simple and practical way to provide ZVS for the power MOSFETs. The peak value of the auxiliary circuit current adaptively changes during the line cycle. The peak value of the auxiliary circuit current is very small near the zero crossing points where a small amount of current is required to guarantee ZVS. On the other hand, the peak value of the auxiliary circuit current is at its maximum for the peak points where the auxiliary circuit current should neutralize the input current. This solution is very practical for automotive applications, since the power circuit can be implemented by the commonly used full-bridge MOSFETs and the only additional components are a small DC-blocking capacitor and a small high frequency inductor. H-bridge power modules are the standard building blocks in automotive applications. 
       FIG. 5  shows the boost inductor valley current, peak current and the envelope of the auxiliary inductor current. In order to guarantee ZVS, the auxiliary inductor current should not only neutralize the valley current, I V , but should also provide enough current to charge and discharge the output capacitors. The valley current, I V , and the peak current, I F , are given by Equations 1 and 2: 
     
       
         
           
             
               
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     Therefore, the envelope of the auxiliary circuit current should be controlled so as to follow the sinusoidal waveform as shown in  FIG. 5  to optimize the amount of reactive current. 
       FIG. 6  shows the block diagram of the control system for the circuit. The control system includes an external voltage loop, an internal current loop, and a switching frequency control loop. The frequency loop is added to the control system to optimize the circulating current of the auxiliary circuit based on the load condition and duty ratio of the converter. As can be seen, the outputs of the control system are signals to be sent to the gate leads of the various boost power transistors of the circuit in  FIG. 2 . The control system therefore controls when, how often, and how long the power transistors in the circuit are activated or are in active mode. 
     At heavy loads, the switching frequency is lower in order to provide more reactive current in the auxiliary circuit to overcome higher values of I V  and to charge and discharge the output capacitors. At light loads, the switching frequency is higher in order to reduce the auxiliary current so that any extra circulating current between the two phases can be avoided. The required auxiliary circuit current for different loads is determined by: 
     
       
         
           
             
               
                 
                   
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     For Equation 4 and subsequent equations, C So  is the combination of the power transistor capacitance and the snubber capacitance. Or, in equation terms, C So =C oss (MOSFET)+C Snubber . 
     The auxiliary circuit current is given by: 
     
       
         
           
             
               
                 
                   
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     The boost inductor ripple is given by: 
     
       
         
           
             
               
                 
                   
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     Inserting Equation 4 and Equation 6 into Equation 5 determines the desired switching frequency of the converter: 
     
       
         
           
             
               
                 
                   
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                   7 
                 
               
             
           
         
       
     
     For the circuit in  FIG. 2 , all the power transistors have the same switching frequency. This switching frequency is determined in two stages. In the first stage, the switching frequency is determined based on the amount of load on the circuit. For the second stage, the switching frequency is modified during the line cycle when the duty ratio becomes lower than 0.5. 
     A 3 KW prototype was implemented to verify the performance of the converter. The converter specifications are shown in Table I and the designed parameters are given in Table II. (see below)  FIG. 7  illustrates the system block diagram. For the main controller, a DSP board from Texas Instruments (eZdSP TMS320F28335) was used. Of course, other types, makes, and models of devices may also be used as the controller. Referring to  FIG. 7 , aat the input stage, there is inrush current protection, which limits the inrush current of the converter. Since there is usually a large capacitor at the output of the PFC, the inrush current to charge the capacitor is very high and a circuit is required to limit this current. The next block is the EMI filter, which is designed to comply with the EMI standard (CISPR25/12) for electric vehicles. The following block is the input diode rectifier. It rectifies the input voltage for the two-phase interleaved boost converter. The interleaved boost converter converts the rectified input voltage to the intermediate DC-bus voltage. The output capacitor of the interleaved boost converter is large (in this implementation the capacitor had a capacitance of 1.4 mF) in order to decrease the 2 nd  harmonic voltage ripple caused by the power ripple of the input boost PFC converter. In addition, there is a differential-Mode (DM) filter at the output of the PFC in order to filter out the differential-mode noise. At the output of this filter, a clean DC-bus voltage is provided to the full-bridge converter. Note that another EMI filter is preferable at the output of the full-bridge converter in order to provide filtering for the EMI noise injected by the inverter. Since the inverter is connected to the high energy battery, it injects switching noise to the battery charger. 
     
       
         
               
             
               
               
               
             
               
               
               
               
             
               
               
               
             
           
               
                 TABLE I 
               
             
             
               
                   
               
               
                 Converter specification. 
               
             
          
           
               
                 Symbol 
                 Parameter 
                 Value 
               
               
                   
               
             
          
           
               
                 Po 
                 Output Power 
                 3  
                 KW 
               
               
                 Vac 
                 Input Voltage 
                 170-267  
                 VAC 
               
               
                 Vo 
                 Output Voltage 
                 235-431  
                 VDC 
               
               
                 f sb   
                 Interleaved 
                 44-148  
                 KHz 
               
               
                   
                 boost switching 
                   
                   
               
               
                   
                 frequency 
                   
                   
               
               
                 f sf   
                 Full-Bridge 
                 220  
                 KHz 
               
             
          
           
               
                   
                 Switching 
                   
               
               
                   
                 Frequency 
                   
               
               
                 I in (max)   
                 Maximum input 
                 16A 
               
               
                   
                 current 
                   
               
               
                 I inrush    
                 Maximum inrush 
                 32A 
               
               
                   
                 current 
                   
               
               
                 P.F. 
                 Power Factor 
                 &gt;98% 
               
               
                   
               
             
          
         
       
     
     
       
         
               
             
               
               
               
             
           
               
                 TABLE II 
               
             
             
               
                   
               
               
                 System parameters. 
               
             
          
           
               
                 Symbol 
                 Parameter 
                 Value 
               
               
                   
               
               
                 L A , L B   
                 Boost Inductors 
                  270 uH 
               
               
                 C o   
                 Output Capacitor 
                  1.4 mF 
               
               
                 S A1 , S A2 ,  
                 MOSFETs 
                 STx25NM50N 
               
               
                 S B1 , S B2   
                   
                   
               
               
                 D BR   
                 Bridge Rectifier 
                 20ETF06pbF 
               
               
                 L AUX   
                 Auxiliary 
                  120 uH 
               
               
                   
                 Inductor 
               
               
                   
               
             
          
         
       
     
       FIG. 8  and  FIG. 9  show the waveforms of the ZVS boost PFC converter. According to these figures, the boost MOSFET is turned-on under zero voltage. This is due to the negative current provided by the auxiliary circuit. Having the MOSFETs at the output side guarantees the appropriate waveforms across the auxiliary circuit and avoids any unwanted turn-on of the output diodes prior to the boost MOSFET turn-on.  FIG. 9  is the enlarged version of  FIG. 8 . This figure shows that the output capacitor of the boost MOSFET is completely discharged prior to applying the gate signal and once the voltage across the MOSFET has become zero the gate signal is applied to the MOSFET. 
       FIG. 10  shows the waveforms of the two phases of the interleaved boost PFC converter as well as the auxiliary circuit current. As can be seen from the figure, the auxiliary circuit provides reactive current that activates the power transistors for both phases at the same time. 
       FIG. 11  illustrates that the auxiliary circuit current changes during a line cycle based on the input current. The auxiliary circuit current is at its minimum at the zero crossing points of the input current and it is at its maximum at the peak of the input current. This implies that the auxiliary circuit current adaptively changes based on the shape of the input current and is optimized over the line cycle. 
       FIG. 12  shows the input voltage and the input current for full-load. It can be seen that the input current and input AC voltage are in phase, thus maintaining a near unity (0.999) power factor. 
       FIG. 13  shows the efficiency curves of a conventional interleaved boost PFC converter along with the efficiency curves for the interleaved boost PFC converter according to one aspect of the invention. From the figure, it can be seen that converter of the invention shows better efficiency for the whole load range compared to the conventional converter circuit. The improvement in the efficiency can be attributed to the fact that the converter of the invention eliminates two main sources of losses, the turn-on losses of the boost MOSFETs and the reverse-recovery losses of the output diodes. 
     It should be noted that while MOSFETs are used in the circuit, other types of transistors may be used. However, MOSFETs are the preferred transistors for this invention as zero voltage switching is more critical and useful to MOSFETs than for other transistors. Also, while the MOSFETs in the converter described above are disclosed as having a separate snubber capacitor, such separate snubber capacitors are preferred but not necessary. The internal capacitance of each MOSFET can serve as the snubber capacitor. 
     The circuit according to one aspect of the invention is also useful for providing ZVS for other types of switches such as IGBTs (insulated gate bipolar transistors). However, other switches may need other types of soft-switching in order to operate efficiently. As an example, IGBTs require ZCS (zero capacitance switching) at turn off to avoid tailing current losses. 
     A person understanding this invention may now conceive of alternative structures and embodiments or variations of the above all of which are intended to fall within the scope of the invention as defined in the claims that follow.

Technology Classification (CPC): 7