Patent Abstract:
Method for nondestructive and noncontact detection of faults in a test piece, with a transmitter coil arrangement with at least one transmitter coil that transmits periodic electromagnetical AC fields to a test piece, a receiver coil arrangement with at least one receiver coil for detecting a periodic electrical signal having a carrier oscillation whose amplitude and/or phase is modulated by a fault in the test piece. A signal processing unit produces a useful signal from the receiver coil signal, and an evaluation unit evaluates the useful signal to detect a fault in the test piece. A self-test unit undertakes systematic quantitative checking of signal processing functions of the signal processing unit and/or of the transmitter coil arrangement and/or of the receiver coil arrangement and/or upon external request undertakes calibration of the signal processing unit using a calibration standard which replaces the transmitter coil arrangement and/or of the receiver coil arrangement.

Full Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     This invention relates to a device and method for nondestructive and noncontact detection of faults in a test piece. In particular, the present invention relates to fault detection using measurements of an eddy current or magnetic flux leakage. Furthermore, the invention relates to a device and a method for detecting electrically conductive particles in a liquid flowing in a pipe segment using the eddy currents induced in the particles being detected. 
     2. Description of Related Art 
     Conventional nondestructive and noncontact detection of faults in a test piece of a semi-finished metallic product is performed by inducing and measuring eddy currents in the test piece. In doing so, the test piece is exposed to periodic alternating electromagnetic fields through a sinusoidally energized transmitter coil. The resulting eddy currents induced in the test piece induce a periodic electrical signal in a coil arrangement which is used as a probe. This periodic electrical signal has a carrier oscillation according to the transmitter carrier frequency whose amplitude and/or phase is modulated by a fault in the test piece when a fault travels into the sensitive region of the probe. Conventionally, when scanning the test piece, the test piece is moved linearly with respect to the probe; however, arrangements with a rotating probe also known. For example, an eddy current measurement device with a linearly advanced test piece is described in U.S. Pat. No. 5,175,498. 
     Similarly, electrically conductive particles in a liquid, which flows through the coils, cause eddy current losses. These eddy currents can be determined by measuring the impedance change of the coils. In this way electrically conductive particles in a liquid flowing in a pipe can be detected by means of an inductive coil arrangement. This is especially advantageous for detection of the concentration of metallic particles in the lubricant circuit of a machine in order to draw conclusions about the machine state such as measurements of machine wear. 
     Another conventional measurement method for nondestructive and noncontact detection of faults in a test piece is magnetic flux leakage measurement (or stray magnetic field measurement), by means of an induction coil with a magnetic yoke, which magnetizes the test piece resulting in a magnetic flux leakage produced by the test piece. The magnetic flux is measured by means of a suitable sensor. Faults in the test piece are detected based on their effects on the magnetic flux leakage. One example of this flux leakage measurement can be found in U.S. Pat. No. 4,445,088. 
     In eddy current measurement devices containing probes which rotate around the periphery of the test piece, measuring the distance between the probe head and test piece is performed in order to correct the measurement with respect to the distance because the distance fluctuates during the course of one revolution. The measurement of the distance is performed because of decentering or asymmetry of the cross section of the test piece occurs during one revolution. One example of this arrangement can be found in German Patent Application No. 40 03 330 A1. 
     International Patent Application Publication WO 2006/007826 A1 discloses an eddy current measurement device with a digital front end, such that the A/D converter stage is triggered with a n-th integral fraction of the frequency of the carrier oscillation, where n is selected depending on the fault frequency, i.e., the quotient of the relative velocity between the test piece and probe and the effective width of the probe. 
     U.S. Pat. No. 4,209,744 describes an eddy current measurement device which has a test means which simulates signals that are typical of faults in a test piece in order to perform fundamental checking of the electronics. However, only a single amplitude and a defined primary fault frequency can be simulated. Even if the simulated fault signal were provided with variations, all the electronics cannot be tested. Furthermore, such a simulated fault signal cannot be traced to a certified reference element without dismounting all the electronics and sending them to a laboratory. 
     International Patent Application Publication WO No. 01/22075 A2 describes an eddy current measurement device within the framework of self-calibration of the system. The intensity of the signal originates from a segment of a test piece which does not contain a fault. 
     GB Patent Application No. 2 192 064 describes an inductive test device where the device is detuned to simulate a fault by a self-test means and by connecting a LED. 
     SUMMARY OF THE INVENTION 
     A primary object of this invention is to devise a device and method for nondestructive and noncontact detection, especially by means of eddy current measurement, or flux leakage measurement, of faults in a test piece or by detecting electrically conductive particles in a liquid flowing in a pipe segment, to ensure that measurement is as reliable as possible. 
     The above object of the invention is achieved in a device as described below. 
     In the approach in accordance with the invention, the self-test unit undertakes systematic quantitative checking of the signal processing functions of the signal processing unit, the transmitting coil arrangement, and the receiver coil arrangement, and upon request to undertake calibration of the signal processing unit with a calibration standard which is to replace the transmitter coil arrangement and/or the receiver coil arrangement. This is advantageous because it allows for comprehensive checking of the functions of the front-end, especially of the filters and amplifiers as well as the probe, and thus, high reliability of the measurement results is achieved. In particular, calibration of the device is also easily enabled. This applies especially to calibration with respect to the adjustable preamplifier. 
     Altogether, increased reliability of the test results is achieved since faults in the individual electronic components of the device can be reliably detected. In particular, high reliability is achieved compared to the calibration known in the prior art on a simulated sample fault since the latter in practice generally does not emerge in the precise form of the simulated fault and thus the meaningfulness of calibration on such a sample fault is relatively low. Further, the individual components cannot be separately quantitatively checked. 
     Instead of using the invention only in the nondestructive and noncontact detection of faults in a moving test piece, i.e., in an eddy current test device or a stray flux measurement device, as described herein, the invention can also be used in the detection of electrically conductive particles in a liquid flowing in a pipe segment with a velocity, such as a particle counter. 
     Preferably, the self-test unit is made to switch the signal processing unit for checking the signal processing functions such that the signal for the transmitter coil arrangement is fed directly as a periodic input signal into the signal processing unit, with the input signal being systematically varied. Typically the signal processing unit has amplifiers and frequency filters, the self-test unit being made to check by means of variation of frequency and amplitude of the signal for the transmitter coil arrangement whether the measured gain of the amplifiers and the measured corner frequencies and steepnesses of the frequency filters are within the given specification, and a corresponding fault signal is output if the specification is not satisfied. 
     Preferably, the driver for the transmitter coil has a current sensor. The self-test unit monitors and determines the impedance of the transmitter coil from the transmitter coil current and the transmitter coil voltage. Preferably, the receiver coil is made in a difference coil arrangement. The self-test unit is determining and monitoring the offset voltage of the receiver coil. Advantageously, the self-test unit is made to store the transmitter coil current and the receiver coil offset voltage as a function of time in order to enable observation of long-term changes of the transmitter coil and the receiver coil. 
     The device can be made with several channels, the transmitter coil arrangement and the receiver coil arrangement have several coils which are each assigned to one certain measurement frequency. 
     Preferably, the calibration standard is at least one RC element, and by means of a calibrated measurement resistance of the RC element the A/D converter, or other converters of the signal processing unit, can be checked with respect to their accuracy, and the sampling frequency of the processor of the signal processing unit can be checked by means of the corner frequency of the RC element. The calibration standard can be a voltage divider which has been certified by a test laboratory. Thus, the sensitivity of the entire system can be checked with a calibrated reference element so that the entire system can be checked at least with a typical setting. 
     Preferably, the front-end is made digital, i.e., the receiver coil signal is sampled by means of a triggerable A/D converter stage and then filtered by means of frequency filters to obtain a demodulated useful signal. The A/D converter stage is triggered with the n-th integral fraction of the frequency of the carrier oscillation of the signal for the transmitter coil arrangement, n is chosen depending on the fault frequency which arises as the quotient of the relative velocity between the test piece and the receiver coil arrangement and the effective width of the receiver coil arrangement, and the frequency filters being set as a function of the fault frequency. 
     Typically, the signal processing unit has an adjustable preamplifier for the receiver coil signal, and the preamplifier can be checked by the calibration standard made as a RC element being exposed to a fixed sinusoidal voltage whose amplitude is chosen such that in the least sensitive setting of the preamplifier a sinusoidal signal can be digitally converted with the desired accuracy by means of the A/D converter stage so that at higher gains of the preamplifier, the sinusoidal signal is overdriven. The overdriven sinusoidal signal is reconstructed with a mathematic approximation, for example, by means of adjustment theory, in order to determine the actual signal amplitude. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a block diagram of an inductive measurement device with a self-test function and calibration function in accordance with the invention; 
         FIG. 2  is a block diagram of an aspect of the invention which is used for detecting faults in a moving test piece; 
         FIG. 3  is a block diagram of one example of an inductive measurement device according to an aspect of the invention which is used for detecting electrically conductive particles in a flowing liquid; 
         FIG. 4  schematically illustrates a longitudinal section through a pipe through which a liquid is flowing and which is provided with a transmitter and receiver coil for use with the measurement device as shown in  FIG. 3 , and 
         FIG. 5  is a block diagram of the wiring of the coils from  FIG. 4 . 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       FIG. 1  shows a block diagram of an inductive measurement device with a self-test function and calibration function according to an aspect of the invention. A signal processor  60  communicates with a programmable array logic (PAL) element  68 . The PAL element  68  is designed to control the A/D converter and D/A converter. The PAL element  68  also supplies a transmitter coil driver  70  which is provided with a current sensor  72 , and delivers the signal for the transmitter coil arrangement (not shown in  FIG. 1 ) of the probe  11  (i.e., measurement head). The receiver coil signal of the receiver coil arrangement (not shown in  FIG. 1 ) of the probe  11  is provided to a low-noise amplifier  74  which is used as a preamplifier. The gain of the low-noise amplifier  74  is controlled or variably set by the processor  60  by way of the PAL element  68 . The signal amplified by the amplifier  74  passes through a resonance filter  78  and is supplied to the PAL element  68  and then the processor  60  for processing, or evaluation, of the signal after digitization in an A/D converter  80 , which can be designed to handle 18 bits. In this way, from the receiver coil signal a usable signal is produced which is then evaluated by an evaluation unit. The evaluation unit can be implemented in the form of the processor  60  and/or externally, for example, as a personal computer (PC)  64 . 
     Furthermore, the system can have a distance sensor  82  with a transmitter coil and a receiver coil (not shown) in order to produce a distance signal from the receiver coil signal of the distance sensor  82 . The distance signal constitutes a measure of the distance between the test piece and the probe  11 . There is a driver  84  for the transmitter coil of the distance sensor  82  which has a current sensor  86  and which is supplied by the PAL element  68 . The receiver coil signal of the distance sensor  82  is supplied to a unit  88  which performs amplification, offset and rectification of the distance signal. The unit  88 , like the amplifier  74 , is controlled by the PAL element  68 . The distance signal is supplied to the PAL element  68 , and then to the processor  60  for evaluation by an A/D converter  90 , which can be designed to handle 16 bits. Further, there can be several distance sensors  82 . 
     The elements  68 ,  70 ,  74 ,  76 ,  78 ,  80 , and optionally  60 , as well as elements  84 ,  86 ,  88 ,  90  are part of the signal processing unit which produces a signal for evaluation by the evaluation unit from the receiver coil signals. 
     A self-test unit  62  is implemented in the processor  60 . The self-test unit  62  performs systematic quantitative checking of the signal processing functions of the signal processing unit of the front end and systematic quantitative checking of the probe  11  and of the distance sensor  82 . Further, the processor  60  performs the checking automatically, at system startup, or at the request of the user interface which can be PC  64  or a touch display  65 . 
     A switch arrangement  66  with three switches  63 ,  67 ,  69  is used for monitoring the signal processing unit. The three switches  63 ,  67 ,  69  can be actuated by the self-test unit  62  (in doing so the switches  63  and  67  are opened and the switch  69  is closed) in order to feed the signal for the transmitter coil of the probe  11  as a periodic input signal into the signal processing unit, i.e., into the input of the amplifier  74  by bypassing the transmitter coil directly. 
     In the self-test, the self-test unit  62  provides the signal for the transmitter coil which is varied with respect to frequency and amplitude in order to check whether the measured gain of the amplifier  74  and the measured corner frequencies and steepness of the frequency filter  78  are within the required specifications. A corresponding fault signal is output to the user interface  64 ,  65  if the specification is not satisfied. 
     According to aspects of the invention, the device can be made with several channels. Transmitter coil driver  70 , the probe  11  and the self-monitoring switch arrangement  66  are provided once for each channel, and a multiplexer  76  is connected upstream of the amplifier  74  (for each transmitter coil there is then its own frequency). 
     A self-test switch arrangement  92  is provided between the driver  84  and the unit  88 . The self-test switch arrangement  92  has three switches  89 ,  91 ,  93  which can be actuated by the self-test unit  62  (in doing so the switches  89  and  91  are opened and the switch  93  is closed) to induce a self-test of the unit  88 , or of the A/D converter  90  by a signal which is output by the transmitter coil driver  84 , and which is bypassing the transmitter coil of the distance sensor  82 . This signal is being sent directly to the input of the unit  88 , and by means of the self test unit  62  the frequency and amplitude of the coil driver signal can be systematically varied. 
     In addition to the output signal of the unit  88 , the current signal of the current sensor  72  and the current signal of the current sensor  86  are supplied to the multiplexer  94 , which is connected upstream of the A/D converter  90 . The sensor current signals are supplied, in this way, to the self-test unit  62  for evaluation. The complex impedance of the respective transmitter coil can be determined and monitored by means of the self-test unit  62  from the transmitter coil current and the transmitter coil voltage detected by current sensors  72  and  86 . Also, a fault signal can be optionally output by way of the user interface  64  and  65 . As illustrated in  FIG. 1 , the transmitter coil voltages are measured at the sites labelled  1  and  3  and are supplied to the PAL element by way of the multiplexer  94  and the A/D converter  90 . 
     Furthermore, the offset voltage of the receiver coil of the probe  11  can be monitored by means of the self-test unit  62  (Note: only difference coils have an offset voltage, which arises in any difference coil arrangement since two coils are never exactly identical). 
     The offset voltage can be eliminated from the receiver signal by means of a high-pass filter. The difference of the voltage before and after the high-pass filter then yields the offset voltage. 
     Advantageously, the self-test unit  62  is made such that the transmitter coil current and the receiver coil offset voltage are stored as a function of time allowing for observation of long-term changes of the transmitter coils and the receiver coils. This monitoring is especially important when the system is designed as an inductive particle counter because the coils cannot be easily dismounted and checked. 
     Furthermore, the self-test unit  62  is configured such that calibration of the signal processing electronics is enabled by means of a certified calibration standard  96  which can replace the coil  11 . The calibration standard  96  is connected on the input side to the transmitter coil driver  70  and on the output side to the multiplexer  76  and to the amplifier  74 . When the calibration standard  96  has several reference elements, such as, different resistances, which are switched over in the course of calibration, the calibration standard  96  has one terminal  98 , for example an I 2 C bus, which is connected to the processor  60  and the self-test unit  62  for undertaking the corresponding switchovers of the reference elements. 
     The points labeled “2” and “4” allow for direct measurement of the voltages upstream of the input channels of the amplifier  74  and of the unit  88 . Thus, it is possible to directly measure the voltage drop with the calibration standard  96  which was set instead of the corresponding coil, for example. 
     It is preferable that the calibration standard has at least one RC element with at least one calibrated measurement resistance for checking the precision of the A/D converter of the signal processing electronics. The sampling frequency of the processor  60  can also be checked with the RC element by using the corner frequency of the RC element which is precisely known. The measurement resistance of the calibration standard  96  is a lowpass filter to suppress interference. As a reference element, the measurement resistance of the calibration standard  96  provides for a defined voltage at the input of the A/D converter  80  so that unwanted fluctuations of the sampling frequency are detected. 
     It is preferable that calibration be performed once a year. 
     The calibration standard  96  may be a separate unit independent of the measurement device and connected to the measurement device only during calibration. This example embodiment is advantageous because the calibration of the calibration standard can be easily checked by a certified calibration laboratory. 
     Alternatively, the calibration standard  96  can be made as a part of the measurement device such as a component provided on a board of the measurement device which is connected in place of the corresponding coil at need. This example embodiment has the advantage that the measurement device does not need to be opened for preparation of calibration. However, in this case, the calibration of the calibration standard cannot be checked. 
     The calibration standard  96  is helpful especially for calibration of the adjustable preamplifier  74 . When the calibration standard  96  for economic reasons has only a single or only a few reference resistance values, it is possible to proceed as follows. The RC element of the calibration standard  96  obtains a fixed sinusoidal voltage from the transmitter coil driver  70 . The fixed sinusoidal voltage is so large that a sinusoidal signal can be digitally converted with a desired accuracy by means of the A/D converter  80  in the least sensitive position of the amplifier  74 . If the gain is increased by means of the PAL element  68 , the sine is cut off at some time, and the truncated sine then can be reconstructed again via mathematical approximation, such as a adjustment theoretical calculation. As a result, the actual amplitude of the signal can be measured. The prerequisite for this method is that the electronics used do not have a latchup effect and the input stage of the A/D converter  80  is protected against destruction by overvoltage. 
     The following equation of the adjustment theoretical computation for a sine may be used:
 
 A 0 *n+A 1*[sin( x )]+ A 2*[cos( x )]=[ yi] 
 
 A 0*[sin( x )]+ A 1*[sin 2 ( x )]+ A 2*[sin( x )*cos( x )]=[ yi *sin( x )]
 
 A 0*[cos( x )]+ A 1*[sin( x )*cos( x )]+ A 2*cos 2 ( s )]=[ yi *cos( x )]
 
where yi is a measured value such that y(i)=A0+A1*sin(x)+A2*cos(x) and x=2*π*f*i*dt, where f indicates the frequency. The brackets stand for summations over the running variable i from zero to n. Those measured values which are outside the allowable range, i.e., the “truncated” values, may not be used here. The value x represents the current angle, which need not be equidistant.
 
     By computing the amount of A1 and A2, the original amplitude A=SQRT(A1*A1+A2*A2) and the phase offset PHI=arctan(A2/A1) are obtained. 
     It goes without saying that the described signal reconstruction can be used not only in the checking of the variable amplifier  74 , but also in an eddy current test, if as a result of certain circumstances the receiver coil signals arise which overdrive the A/D converter. Ultimately, the measurement range can be expanded by this signal reconstruction using only software. 
     The relatively simple checking of the variable amplifier  74  described above allows for the storage and use of correction values for the respective gain, allowing for more economical amplifiers of the same quality. 
     There are resonance filters, like the resonance filter (or a combination of highpass and lowpass)  78 , allowing for operation with a variable transmitter frequency. The most favorable sampling frequency arising as a function of the velocity of the test piece, effective coil width, and transmitting frequency. As already described, in a self-test using variation of the frequency and amplitude of the input voltage, the corner frequencies and the edge steepness of the filters can be determined. 
     Changes of the sensor hardware, especially damage, can be ascertained early by the described impedance measurements of the transmitter and receiver windings using the self-test unit  62  so that test times with damaged sensor hardware can be avoided as much as possible. As a result, measurement becomes more reliable. 
     The described measurement of the receiver coil offset voltage by the self-test unit  62  enables early detection of overdriving problems, for example, in conjunction with certain test piece materials. This allows for preventive reactions to problems and increases in the reliability of the test. 
     The possibility of calibration of the system by means of the self-test unit  62  and the calibration standard  96  enables simple calibration of the system on site, eliminating the necessity of installation and dismounting of a test adapter in the system. As a result, production and servicing of the system is more economical, because an adaptation of a front end in a testing device is not needed. 
     The calibration standard  96 , itself, if it is made as a separate unit, can of course also be calibrated at regular intervals by a certified calibration laboratory, as previously described. 
       FIG. 2  illustrates a block diagram of an example of an inductive measurement device according to an aspect of the claimed invention which is used for detecting faults in a moving test piece and a digital demodulation method. Aside from the self-test function and signal reconstruction, this device is described in WO 2006/007826 A1. Here, a test piece  13  in the form of a semi-finished industrial article, for example, a slab, which is tested when it moves linearly with a variable velocity v past the probe  11 . The velocity is detected with a velocity detector  21  which can deliver for example a signal essentially proportional to the velocity v. The signal can be, for example, a rectangular signal (possibly also two-track in order to be able to distinguish forward and backward) which contains one pulse, for example, per 5 mm advance of the test piece  13 . 
     The probe  11  has a transmitter in the form of a transmitter coil  18  and a receiving coil  15 . With an alternating electromagnetic field with at least one given carrier frequency, the transmitter coil  18  is used to induce eddy currents in the test piece  13 . These eddy currents in turn induce an AC voltage in the receiving coil  15  which AC voltage acts as a probe signal and has a carrier oscillation with the carrier frequency of the transmitter coil  18 . The amplitude and the phase of the probe signal is modulated by a fault  23  when the fault  23  travels into the effective width WB of the receiving coil  15 . The receiving coil  15  is preferably made as a difference coil, i.e., a coil with two windings which are wound in the opposite direction, and react only to changes of the electrical properties due to the presence of a fault  23  of the test piece. Difference coils are suitable mainly for detection of sudden changes in the test piece  13 . An absolute coil can also be used as the receiving coil  15  which comprises several windings wound in the same direction, and suitable especially for detection of long homogeneous changes in the test piece  13 . 
     The voltage for the transmitter coil  18  can be produced by a binary signal produced by a timer unit  44  and delivered to a generator  48  as the input frequency which produces a rectangular signal or a sinusoidal signal which travels through the curve shaper  40  and then is amplified by a power amplifier  42  before being sent to the transmitter coil  18 . Preferably, the signal has a sinusoidal shape and in the simplest case contains only a single carrier frequency, but measurements with several carrier frequencies at the same time and/or carrier signals which differ distinctly from sinusoidal oscillations are also possible. Typically the carrier frequency is in the range from 1 kHz to 5 MHz. 
     Fundamentally, the transmitter coil can also be operated with a digital signal based on the pulse duration modulation. This has the advantage of greatly reducing the power loss in the driver stage. 
     The probe signal received by the receiving coil  15  travels through a bandpass filter  19  and an adjustable preamplifier  17  before being supplied to an A/D converter stage  35 . The bandpass filter  19  is used, on the one hand, as an (anti-)aliasing filter with respect to signal digitization by the A/D converter stage  35 , and on the other hand, to mask out high frequency and low frequency noise signals. The adjustable preamplifier  17  is used to bring the amplitude of the analog probe signal to the amplitude optimally suitable for A/D converter stage  35 . 
     The A/D converter stage  35  has two A/D converters  32  and  34  which are connected in parallel and have high resolution with a resolution of at least 16 bits, preferably at least 22 bits. It is also preferable that the A/D converter stage  35  is able to carry out at least 500 A/D conversions per second. The A/D converters  32 ,  34  are preferably flash converters or SAR (successive approximation register) converters. 
     The version with two A/D converters is one example. It is important that the fault signal is orthogonally sampled, which may also be performed with only one converter. 
     The A/D converter stage  35  is triggered by a trigger means  37 , which has the aforementioned timer unit  44 , a cosine generator  48 , a sine generator  46  located parallel to the cosine generator  48 , and a frequency divider  30 . The signal which has been generated by the cosine generator  48  and which has the frequency of the carrier frequency of the supply signal of the transmitter coil  18  is provided to the frequency divider  30 . The signal of the sine generator  46  which corresponds to the signal of the cosine generator  48 , but with a phase-shift of 90° thereto, is also provided to frequency divider  30 . In the frequency divider  30  these two signals are divided with respect to their frequency by a whole number n. The corresponding frequency-reduced output signal is used to trigger the A/D converter  32  and the A/D converter  34 . The selection of the number n for the divider  30  is undertaken by a digital signal processor  60  depending on the fault frequency, i.e., the quotient of the current velocity of the test piece v and the effective width WB of the receiving coil  15 . Preferably, n is chosen to be inversely proportional to the main fault frequency in order for the trigger rate of the A/D converter stage  35  to be at least roughly proportional to the main fault frequency. This results in that if the effective width WB in the first approximation is assumed to be constant, at a higher test piece velocity v and thus a high fault frequency the analog probe signal is sampled more frequently. 
     Preferably, the divider  30  is made as a so-called PAL (Programmable Array Logic) component in order to ensure that the trigger signals arrive synchronously, to the output signal of the cosine generator  48  and the sine generator  46  without phase jitter at the A/D converter stage  35 . 
     Due to the corresponding phase shift of the two input signals of the divider  30 , the two A/D converters  32 ,  34  are also triggered with a fixed phase offset of 90°. In this way the analog probe signal can be evaluated in a two-component manner, i.e., with respect to amplitude and phase. It goes without saying that the phase delay between the trigger signal of the A/D converter signal  35  and the signal of the transmitter coil  18  should be as small as possible, and especially so-called phase jitter should also be avoided, i.e., the phase relations should be constant in time as exactly as possible. 
     With the illustrated trigger means  37  the analog probe signal is sampled by each A/D converter  32 ,  34  at most once per full wave of the carrier oscillation (in this case n is equal to 1). Depending on the current fault frequency, i.e., the velocity of the test piece v, n can be much larger than 1 so that sampling is performed only in each n-th full wave of the carrier oscillation. 
     As already mentioned, what matters is that sampling is taken orthogonally. When sampling is done at 0° and 90° the complex components of the fault signal are obtained. At 180° and 270° the same components are obtained, but in the inverse to those taken at 0° and 90°. By inverting these components an average can be formed and thus an increased sampling rate can be used. Such sampling methods have advantages with respect to noise and design of the input filter. 
     The demodulated, digital, two-channel output signal of the A/D converter stage  35  travels through a digital bandpass filter  52  which can be the signal processor  60 . The digital bandpass filter  52  is used to mask out noise signals outside the bandwidth of the fault signal. For this purpose, the corner frequency of the highpass filter (software filter) is preferably chosen such that it is less than one fourth of the fault frequency, while the corner frequency of the lowpass filter is preferably chosen such that it is at least twice the fault frequency to avoid masking out the signal portions which still contain information of the fault. 
     The digital bandpass  52  is clocked with the sampling rate of the A/D converter stage  35 , i.e., the trigger rate. This has the advantage that the corner frequencies of the bandpass are automatically entrained with the fault frequency when the fault frequency changes, i.e., when the velocity of the test piece v changes, since the corner frequencies of a digital bandpass filter are proportional to the clock rate and the clock rate is automatically adapted to the change of the fault frequency by way of the sampling rate which is stipulated by the trigger unit  37 . 
     This also applies analogously when the transmission frequency has been changed. This reduces the cost of digital filtration with respect to different types of filter stages. 
     The information necessary for determining the main fault frequency with respect to the effective width WB can be either manually input to the signal processor  60  made available directly by the probe  11 , as described, for example, in European Patent Application No. 0 734 522 B1. 
     It goes without saying that the measurement system reacts analogously to the change of the fault frequency which is caused when the velocity v of the test piece remains constant, but the receiving coil  15  is replaced by another with a different effective width WB. 
     The useful signal, which is obtained after filtration by the digital bandpass filter  52 , is evaluated in a known manner by an evaluation unit  50  in order to detect and locate faults  23  of the test piece  13 . For detection both the amplitude information and the phase information of the fault signal is used. 
     In particular, for relatively large values of n, i.e., when only a relatively small number of full waves of the carrier oscillation are sampled, the transmitter coil  15  and/or the evaluation electronics, especially the signal processor  60 , can be turned off or put on stand-by in order to reduce power consumption during the sampling pauses. Such capability is important especially for portable measurement devices. 
     In the processor  60  the self-test unit  62  for the monitoring and calibration functions named above in conjunction with  FIG. 1  is implemented. Thus, the self-test unit  62  controls the switch arrangement  66  with three switches  63 ,  67 ,  69  in order to feed the signal for the transmitter coil  18  of the probe  11  bypassing the transmitter coil  18  and the receiver coil  15  directly as a periodic input signal into the signal processing, i.e., into the input of the bandpass filter  19 . 
       FIGS. 3 to 5  show one example of an inductive measurement device according to an aspect of the claimed invention used to detect electrically conductive particles in a flowing liquid using a digital demodulation method. Aside from the self-test function, this device is described in the German patent application not published beforehand, with application number of 10 2007 039 434.0 and corresponding to U.S. Patent Application Publication No. 2009/0051350. Fundamentally, the signal processing, especially the signal reconstruction when the A/D converter is being overdriven, and the self-test functions are performed analogously to the above described approach shown in  FIG. 2 . 
     As shown in  FIG. 4 , a pipe segment  10  is surrounded by a first inductive receiver coil  12  and a second inductive receiver coil  14  which is spaced apart from the receiver coil  12  in the axial direction so that a liquid  16  flowing in the pipe segment  10  flows through the coils  12  and  14  in the axial direction. The axial distance of the two coils  12 ,  14  and the axial dimensions of the coils  12 ,  14  are, for example, 2 mm. The two receiver coils  12 ,  14  are surrounded on the outside by a transmitting coil  18  which is located coaxially to the two coils  12 ,  14  and has a larger diameter than coils  12 ,  14 . The axial dimensioning of the transmitter coil  18  is such that the two receiver coils  12 ,  14  are located completely within the transmitter coil  18 . Preferably the extension of the transmitter coil  18  in the axial direction is at least twice as great as the axial extension of the arrangement of the receiver coils  12 ,  14 , i.e., the distance plus the axial extension of the coils  12 ,  14 . The coils  12 ,  14 ,  18  are located in a housing  22  which surrounds the pipe segment  10  and form a probe  11 . 
     Typically, the pipe segment  10  is part of the lubricant circuit of a machine, then the liquid  16 , for example, is a lubricant containing metal particles which are typically abrasion from moving parts of the machine. A typical value for the lubricant flow rate in the main flow is 10 liters/min. At much higher flow rates it is advantageous to measure a secondary flow, instead of the main flow. 
     As shown in  FIG. 5 , the two receiver coils  12 ,  14  are connected subtractively as a difference coil  15 , i.e., they are wound in opposite directions, so that a voltage with the same amount but with opposite signs is induced in the two coils  12 ,  14 . The transmitter coils  18  and the receiver coils  12 ,  14  form a transformer arrangement, where the transmitter coil  18  forms the primary side and the receiver coils  12 ,  14  form the secondary side. The transformer core is formed by the materials or media fed through the coils  12 ,  14 ,  18 , e.g., air, the housing  22 , the pipe  10 , and the liquid  16  with the particles  20 . 
     The impedance difference of the coils  12 ,  14  caused by the particles  20 , i.e. the difference of the impedance of the two coils  12 ,  14  caused by the instantaneous presence of a particle  20  in one of the two coils  12 ,  14  (the particles  20  are much smaller than the distance of the coils  12 ,  14 ), is imaged by the measurement signal output by the coils  12  and  14 . 
       FIG. 3  shows one example of the structure of the eddy current measurement device that uses the probe  11  according to an aspect of the present invention. 
     The transmitter coil  18  is used, by means of an alternating electromagnetic field with at least one given carrier frequency, to induce eddy currents in the particles  20 , which in turn induce an AC voltage that acts as the probe signal in the receiving coil  15 , which is a difference coil. The induced AV voltage in the receiver coil has a carrier oscillation with the carrier frequency of the transmitter coil  18 . The amplitude and the phase of the probe signal are modulated by a particle  20  when the latter travels into the effective width WB of the receiving coil  15 . 
     The voltage of the transmitter coil  18  can be produced, for example, by a binary signal produced by a timer unit  44  input to a generator  48  producing a rectangular signal or a sinusoidal signal, which travels through the curve shaper  40  and then is amplified by a power amplifier  42  before being sent to the transmitter coil  18 . Preferably the signal has a sinusoidal shape and in the simplest case contains only a single carrier frequency, but may also contain several carrier frequencies at the same time and/or carrier signals which differ distinctly from sinusoidal oscillations. Typically the carrier frequency is in the range from 5 kHz to 1 MHz. 
     The probe signal received by the receiving coil  15  travels through a bandpass filter  19  and an adjustable preamplifier  17  before being supplied to an A/D converter stage  35 . The bandpass filter  19  is used, on the one hand, by means of a lowpass filter as an (anti-)aliasing filter with respect to signal digitization by the A/D converter stage  35 , and on the other hand, by means of a highpass to mask out low frequency noise signals. The adjustable preamplifier  17  is used to bring the amplitude of the analog probe signal to the amplitude optimally suitable for the A/D converter stage  35 . 
     The A/D converter stage  35  has two A/D converters  32 ,  34  which are connected in parallel and have high resolution with a resolution of at least 16 bits, preferably at least 22 bits, and are able to carry out at least 500 A/D conversions per second. The A/D converters  32 ,  34  are preferably made as flash converters or SAR (successive approximation register) converters. 
     If offset voltage compensation takes place by means of an additional D/A converter and subtractor, a resolution of the A/D converter of 12 bits is sufficient. 
     The A/D converter stage  35  is triggered by a trigger means  37  which has the aforementioned timer unit  44 , the cosine generator  48 , the sine generator  46  which is located parallel to the cosine generator  48 , and the frequency divider  30 . A signal is provided to the frequency divider  30 . The signal has been generated by the cosine generator  48  and has the frequency of the carrier frequency of the supply signal of the transmitter coil  18 , and the signal of the sine generator  46  which corresponds to the signal of the cosine generator  48 , but which is phase-shifted by 90° to with respect to the signal of the cosine generator  48 . In the frequency divider  30  these two signals are divided with respect to their frequency by a whole number n. The corresponding frequency-reduced output signal is used to trigger the A/D converter  32  and the A/D converter  34 . The selection of the number n for the divider  30  is undertaken by a digital signal processor  60  depending on the particle frequency, which is the quotient of the flow velocity v of the liquid  16 , i.e. the velocity v of the particles  20 , and the effective width WB of the receiving coil  15 . Preferably, n is chosen to be inversely proportional to the particle frequency in order for the trigger rate of the A/D converter stage  35  to be at least roughly proportional to the particle frequency. Therefore, if the effective width WB in the first approximation is assumed to be constant, at a higher flow/particle velocity v and thus higher particle frequency the analog probe signal is sampled more frequently. 
     Preferably, the divider  30  is made as a so-called PAL (Programmable Array Logic) component in order to ensure that the trigger signals arrive with minimum delay i.e. as synchronously as possible with the output signal of the cosine generator  48  and the sine generator  46  and without phase jitter at the A/D converter stage  35 . 
     Due to the corresponding phase shift of the two input signals of the divider  30 , the two A/D converters  32 ,  34  are also triggered with a fixed phase offset of 90°. In this way, the analog probe signal can be evaluated in a two-component manner, i.e., both with respect to amplitude and phase. It goes without saying that the phase delay between the trigger signal of the A/D converter signal  35  and the signal of the transmitter coil  18  should be as small as possible, and especially so-called phase jitter should also be avoided, i.e., the phase relations should be as constant in time as possible. 
     It is ensured that the analog probe signal is sampled by each A/D converter  32  and  34  at most once per full wave of the carrier oscillation (in this case n is equal to 1) with the illustrated trigger means  37 . Depending on the current particle frequency, i.e., the velocity of the liquid v, n however can be much greater than 1 so that sampling only takes place at each n-th full wave of the carrier oscillation. 
     Since sampling takes place at most once per full wave per A/D converter  32 ,  34 , the frequency of the carrier oscillation, i.e., the carrier frequency, is eliminated from the digital signal by this undersampling, i.e., demodulation of the analog probe signal takes place by means of undersampling. 
     Preferably, n is chosen such that a noticeable particle signal is observed in the time interval. That is, a time interval is chosen such that one point of a particle  20  moves through the effective width WB of the receiving coil  15  in this time interval which corresponds essentially to the inverse of the main particle frequency, which is at least 5, preferably at least 20 samples are taken by each A/D converter  32  and  34  to obtain enough information contained in the particle signal sufficient for reliable particle detection. Generally however not more than 50, at most 100, samplings will be necessary during this time interval, a minimum of 10 samplings. 
     The frequency of the carrier oscillation should be chosen such that it is at least ten times the particle frequency, since otherwise the particle signal is carried by too few full waves of the carrier oscillation and the reproducibility of particle detection becomes a problem. 
     The demodulated, digital, two-channel output signal of the A/D converter stage  35  travels through a digital bandpass filter  52  which may be the signal processor  60  and which is used to mask out noise signals which are outside the bandwidth of the particle signal. For this purpose, the corner frequency of the highpass is preferably chosen such that the corner frequency is less than one fourth of the particle frequency, while the corner frequency of the lowpass filter is preferably chosen such that it is at least twice the particle frequency in order to avoid masking out the signal portions which still contain information with respect to particle passage. 
     The digital bandpass filter  52  is clocked with the sampling rate of the A/D converter stage  35 , i.e., the trigger rate; this entails the major advantage that the corner frequencies of the bandpass filter when the particle frequency changes, i.e., when the velocity of the particles v changes, are automatically entrained with the particle frequency since the corner frequencies of a digital bandpass filter are proportional to the clock rate which is automatically adapted to the change of the particle frequency by way of the sampling rate which is stipulated by the trigger unit  37 . 
     The information which is necessary for determining the main particle frequency with respect to the effective width WB can be either input manually to the signal processor  60  or provided directly by the measurement head  11 , as is described for example in European Patent Application No. 0 734 522 B1 and corresponding to International Patent Application Publication. No. 95/16912. 
     It goes without saying that the measurement system reacts analogously to the change of the particle frequency which is caused when the particle velocity v is kept constant, but the receiving coil  15  is replaced by another with a different effective width WB. 
     In particular, for relatively large values of n, i.e., when only a relatively small number of full waves of the carrier oscillation at all is sampled, for example the transmitter coil  18  and/or the evaluation electronics, i.e., especially the signal processor  60 , can be turned off or put on stand-by during the sampling pauses in order to reduce power consumption. This is important especially for portable measurement devices. 
     The useful signal obtained after filtration by the digital bandpass filter  52  is evaluated in an evaluation unit  50  in order to detect the passage of particles  20  using the amplitude and phase information of the particle signal. 
     Advantageously, the evaluation unit  50  is made such that the detected particle passages are counted so that conclusions can be made about the particle concentration in the liquid  16 , and the state of the machine. 
     Fundamentally, in a difference coil, as a result of difference formation (the individual coils of the difference coil are never exactly alike in practice), the so-called coil offset voltage arises that can exceed the actual fault signal by several orders of amplitude, for example, by 100 to 30000 times. The resulting relatively large amplitude of the receiver coil signal compared to the actual useful signal imposes high demands on the electronics, especially on the resolution of the A/D converter. 
     Monitoring and calibration functions, which are named above in conjunction with  FIG. 1 , are implemented in the self-test unit  62  of the processor  60 . Thus, the self-test unit  62  controls the switch arrangement  66  with three switches  63 ,  67 ,  69  in order to feed the signal for the transmitter coil  18  of the probe  11  by bypassing the transmitter coil  18  and the receiver coil  15  directly as a periodic input signal into the signal processing, i.e., into the input of the bandpass filter  19 .

Technology Classification (CPC): 6