Patent Abstract:
This invention describes how to quickly cancel DC offsets that are present in the two quadrature paths of a zero intermediate frequency transceiver. Previously known techniques are not suitable for the 5 GHz WLAN standards because of the very short transmit to receive turn around times and extraordinarily large dc offsets in these systems. This invention solves the above problems. The present invention uses both AC and DC coupling along with automatic gain control techniques to remove unwanted DC offsets within an acceptable time period. The invention further uses a digital signal processor to estimate and subtract out the DC offset errors using time averaged signals. The digital signal processing circuit is capable of further AC filtering and Analog to Digital conversions.

Full Description:
BACKGROUND OF INVENTION 
   The invention relates to a method and apparatus for correcting DC offset problems found in high-speed transceiver systems. 
   Conventional transceiver systems commonly must switch back and forth between transmit and receive modes. As the speed of data transmission increases, the time allowed for switching between the Transmit (Tx) and Receive (Rx) modes becomes smaller. One conventional type of receiver is known as a Direct Conversion zero intermediate frequency (IF) receiver. In this type of receiver a local oscillator is tuned to the carrier frequency of the incoming signal. These types of receivers commonly have multiple stages in which the incoming signal is down converted and processed using a local oscillator (LO) circuit. These IF type receivers create in phase (I) and 90 degrees out of phase quadrature (Q) signals from the received signal. Large DC offsets are produced in the down converter outputs of Zero IF receivers due to LO leakage at the RF ports of the down converters. Additional DC offsets exist along the I and Q paths that include low-pass channel filters and automatic gain control (AGC) circuits with large gains. Therefore each of these DC coupled stages introduces a DC offset error into the signal. In an orthogonal frequency division multiplexing (OFDM) system, a difference between the local oscillator (LO) frequency and the incoming signal frequency causes DC offset errors within the system to profoundly degrade the SNR after demodulation. These unwanted DC errors may also cause the amplifiers used in the I and Q branches to saturate. Once the amplifier is in a saturated state, the received data signal cannot be processed and amplified correctly so the received data signal is lost. 
   Prior art attempts to deal with the above problems have been only semi-successful. Stroet et al.&#39;s article entitled “A Zero-IF Single Chip Transceiver for up to 22 Mb/s QPSK 802.11b Wireless LAN” shows that these offset errors may be reduced or settled in 25 microseconds. The reduced DC offset is still too high for OFDM systems and takes too long to settle. As mentioned above, with an increase in data speeds, this prior art system is not useable in today&#39;s transceiver environments. 
   All known prior art techniques used to reduce these DC offset errors have drawbacks in one form or another. For example, AC coupling signals with high frequency cut off values may reduce response time, but has an unacceptable signal to noise ratio or an unacceptable effect on the signal itself Further, automatic gain control is also only useable when the DC offset level is very small. The end result is that these DC offset values can not be reduced in an acceptable amount of time. 
   SUMMARY OF THE INVENTION 
   The instant invention uses an automatic gain controller AGC and a digital signal processor along with zero IF transceiver circuitry to create I and Q signals to process and remove DC offset signals in acceptable time periods. Using a combination of techniques such as AC and DC coupling, automatic gain control, and digital signal processing, the DC offsets are removed to insignificant levels. The main features or steps of the invention are: 1) dynamically changing the cut off frequency of an AC coupling stage; 2) computing a DC signal error over a time period in which the I and Q signals complete a single or multiple cycles, 3) subtracting the estimated DC errors from the I and Q signals, and 4) high-pass filtering the resultant signal so that residual DC errors are removed. The implementation and timing of these steps allows for DC offset control that is far superior to prior art systems. 
   In a second embodiment a D/A converter is used in each of the I and Q branches to significantly reduce the DC offset, and then steps  2 – 4  as mentioned above are followed. The DSP does a coarse DC correction with the D/A converter, and maintains a list of correction values for all combinations of antenna diversity, LNA amplifier gains and AGC gains. This ensures that DC offset correction values are maintained for all combinations of antenna diversity and low noise amplifier gains. In the second embodiment, the DSP controls the system so that the DC offset error is reduced when AGC gain is reduced. 
   In a third embodiment, the transceiver frequency error is further removed in the receiver by changing the voltage controlled oscillator crystal (VCXO) frequency that is used for the local oscillator (LO) frequency synthesizer. This also ensures that the effect of the DC offset error is minimized after demodulating an OFDM signal. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  shows the transceiver circuit of a first and second embodiment of the present invention. 
       FIG. 2  shows a timing diagram of the present invention. 
       FIG. 3  shows a frequency adjust circuit employed in a third embodiment of the invention. 
       FIG. 4  shows one embodiment of an automatic gain control circuit of the present invention. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   Using a combination of techniques such as AC and DC coupling, automatic gain control, and digital signal processing, the DC offsets are removed to insignificant levels. The main features or steps of the invention are dynamically changing the cut off frequency of an AC coupling stage; computing a DC signal error over a time period in which the I and Q signals complete a single or multiple cycles, subtracting the estimated DC errors from the I and Q signals, and high-pass filtering the resultant signal so that residual DC errors are removed. The implementation and timing of these steps allows for DC offset control that is far superior to prior art systems. 
   In a second embodiment a D/A converter is used in each of the I and Q branches to significantly reduce the DC offset, and the steps as mentioned above are followed. The DSP does a course DC correction with the D/A converter, and maintains a list of correction values for all combinations of antenna diversity and LNA amplifier gains. This ensures that DC offset correction values are maintained for all combinations of antenna diversity and low noise amplifier gains. In the second embodiment, the DSP controls the system so that the DC offset error is reduced when AGC gain is reduced. 
   In a third embodiment, the transceiver frequency error is further removed in the receiver by changing the voltage controlled oscillator crystal (VCXO) frequency that is used for the local oscillator (LO) frequency synthesizer. This also ensures that the effect of the DC offset error is minimized after demodulating an OFDM signal. 
     FIG. 1  shows a diagram of the transceiver  10  as a zero intermediate frequency radio device according to the present invention. The transceiver  10  comprises a receive branch Rx and a transmit branch Tx. A transmit power amplifier  14  is coupled to a Tx/Rx switch  13 . The Tx/Rx switch  13  is coupled to an antenna  11 . The transmitting branch is well known in the art and is not shown in detail here. The receiver branch further includes a variable gain low noise radio frequency amplifier (LNA)  15  that is coupled to the Tx/Rx switch  13 . The LNA  15  amplifies an input signal that corresponds to an incoming radio frequency signal that is received by the antenna  11 . 
   The output of the LNA  15  is coupled to a frequency down converter  34  for down converting the radio frequency signal to a zero intermediate frequency (IF) signal. The present invention employs a quadrature frequency down converter. The frequency down converter  34  which contains mixers  16  and  17  in respective quadrature and in-phase mixer paths that provide filtered and amplified quadrature signals Q and I. The frequency down converter  34  further includes controllable AC couplers  22  and  23 , and channel filters  24  and  25 . The AC couplers  22  and  23  are coupled between the mixers  16  and  17  and the zero IF amplifiers filters  24  and  25 . 
   Control signals for the automatic gain control AGC  31  are provided by a DSP baseband processor  30 . The baseband processor  30  contains processing means for providing cut-off frequency control signals and signals to the AGC unit  31 . Signal line  35  controls the automatic gain controlling data, while signal line  36  contains frequency cutoff information. Also shown is a State Machine  9 . The State Machine  9  is connected to capacitors  22 – 23  and  32 – 33 . The State Machine is used to change the AC coupling frequency from 10 MHz to 500 KHz as will be explained in greater detail below. The operation and functions of the DSP  30  will also be described in detail with reference to  FIG. 2  below. 
   In another embodiment, the baseband circuit  30  further employs analog to digital converters  20  and  21  for canceling the DC offset in the quadrature signals I and Q. The sampled I and Q DC correction signals are supplied by the digital signal processor (DSP)  30 . 
   The transceiver  10  further comprises a PLL  19  for generating local oscillator signals for the receive branch Rx and for the transmit branch Tx. As is well known in the art, the PLL comprises a voltage controlled oscillator (VCO), a loop filter, and an integrator. A reference oscillator signal, as shown in more detail in  FIG. 3 , is supplied to the PLL. In order to generate the I and Q signals, a ninety degrees phase shifter is used in conjunction with the LO signals that are fed to the mixers  16  and  17 . The transceiver  10  does not use the D/A converters in the first embodiment but does employ them in a second embodiment. 
   As described above, the problems with prior art transceiver systems are the unwanted DC offset values that are produced by the system components. 
     FIG. 2  shows a timing diagram according to one preferred embodiment of the invention. This timing diagram shows the durations of the AC and DC coupling stages necessary to reduce the DC offset errors. Also shown in  FIG. 2  are the lower cut off frequencies of the filters during these AC coupling stages. 
   The invention implements both AC and DC coupling in the receiver I and Q base band paths. Temporary AC coupling is used to remove DC offsets that could otherwise saturate the receiver outputs due to the large gain in the base band paths from the down converters outputs to the I and Q outputs. The AC coupling is implemented as a cascade of one or more first order high pass filters  24  and  25 , with a particular lower 3 dB cut off frequency (f lower). 
   With reference to  FIG. 2 , upon entering the receive (RX) mode at time zero, (step  1 ) the AC coupling cutoff frequency (f lower) is momentarily kept at 10 Mz for 0.15 usec. This is done automatically by a state machine  9  in the receiver. This quickly removes all the DC offset in the receiver I and Q base band paths. After 0.15 microseconds (f lower) is automatically reduced to 500 kHz, and remains at this cut off value until the DSP removes the AC coupling and introduces DC coupling. With the 500 kHz AC coupling, DC offset changes (with AGC gain changes) are quickly removed before the signal is sampled or the next AGC iteration. It is during this stage of step  1  that automatic gain control is performed on the signal. The gains of amplifiers  26 , 27 , 28  and  29  are changed by the DSP to adjust the IQ signals to a desired level at the AID input.  FIG. 2  shows 3 distinct time periods of adjustment, however more periods could be used as necessary. In the example shown in  FIG. 2 , the DC coupling (step  2 ) is switched on at 1.05 microseconds. 
   The DC coupling is actually an AC coupling with a very low value of (f lower) which is less than 100 Hz. With such a low cut off frequency, it may be considered DC coupling even for long IEEE802.11a data packets that may be up to 5–6 milliseconds in duration. 
   It should be noted that whenever the LNA  15  gain is changed, there is a change in logic level at the antenna select input, and the state machine  9  changes (f lower) to 10 MHz for 0.15 microseconds and then returns (f lower) to 500 kHz. 
   It should also be noted that when the state machine  9  changes (f lower) from 10 MHz to 500 kHz, there is a step in the I and Q DC levels that may be as large as the peak signal level. This step quickly decays away to very low levels within about 0.8 microseconds. Simulations show that with 500 kHz AC coupling for IEEE802.11a, if a moving average signal power estimate is computed by the DSP  30  (averaged over 0.8 microseconds), then the error is within 2 dB after the first 0.8 microseconds of the RF burst. For a coarse signal level estimate, a 150 nanosecond averaging window (samples at 40 or 80 MHz AID) is sufficient. 
   When the DSP  30  changes the receiver base band AGC gain in step  1 , it must wait for up to 300 nanoseconds before sampling the I and Q signals for computing the signal power. This is because of the transient settling of the receiver AGC DC levels that takes less than 300 nanoseconds to settle for a 500 kHz AC coupling. For small changes in AGC gain settings, the transient settling time is less than 300 nanoseconds. All changes in LNA and AGC gain settings should be done in order to get the proper signal level and allow the DC errors to be removed before making the decision for the next gain setting. 
   After finally adjusting the receiver AGC gain, the DSP  30  changes the receiver paths to DC coupling (step  2 ). When this is done, there is a small change in the I and Q DC levels, and it can not be avoided. It is less than about −5 dB relative to the peak signal level and is due to the AC coupling acting on the signal itself (not related to the actual DC error in the circuit). This DC error remains nearly constant during the rest of the receive burst that may be up to 6 ms long. In this preferred embodiment, this static DC error should be removed digitally by the DSP  30 , only after the I and Q A/D conversion takes place. This will ensure not to degrade the signal to noise ratio (SNR) after the fast Fourier transform (FFT) in the receiver, especially when there are large relative frequency offsets between the Transmitting and Receiving modes in the transceivers. 
   In step  3  of the preferred embodiment the DSP  30  computes the DC offsets in the I and Q remaining parts of the signal. The average values of the I and Q signals are calculated by the DSP  30 . The computed average DC offsets should then be subtracted from their respective signals for the rest of the packet. It is important to calculate the DC offset error at this point while the signal is DC coupled, so as to get an accurate indication of the offset error. After this subtraction, a first order high pass filtering should be done on the following I and Q signals digitally by the DSP  30 , which represents step  4  of the preferred embodiment. This process significantly improves the SNR of the signal. This is because the estimated DC offset is not the true DC offset when the transmitter-receiver frequency offset is present. A residual DC error remains and it must be removed. Simple high pass digital filtering in the DSP  30  is sufficient. A lower cutoff frequency of 1 kHz is optimum for this digital filtering. 
   This process as shown in  FIG. 2  allows the DC offset error to be reduced to acceptable levels within 8 microseconds. In this preferred embodiment it is assumed that the DSP  30  takes 150 nsec to compute the signal power and program the AGC. Actual time for worst case AGC setting will depend on the exact processing delay of the DSP, and the total number of AGC set iterations, and whether or not antenna diversity is used. 
   In another preferred embodiment, instead of computing the DC error in the DSP and subtracting it, the DSP can instead ramp down the value of (f lower) the AC coupling cut off frequency, from 500 kHz to less than 100 Hz over about 4 microseconds. This avoids the sudden step in DC offset that is associated with abruptly changing the cut off frequency. 
     FIG. 3  shows a third embodiment of the present invention, wherein a frequency adjust signal is applied to the voltage controlled crystal oscillator  50  to change the frequency of the LO. A signal (F adjust) is applied to the voltage controlled crystal  50  from the DSP controller. The phase locked loop contains conventional components such as a charge pump multiplier  51 , a low pass filter  52 , an integrator  53  and voltage controlled oscillator  54 . The function of this circuit shown in  FIG. 3  is to ensure that the frequencies of the local and received carriers are the same. When the frequency error is significantly reduced, even a large DC offset does not degrade the SNR after demodulation of the OFDM signal. Therefore the DC estimation and subtraction, and removal of residual DC error, is not required. 
     FIG. 4  shows one embodiment of how the automatic gain control circuit (AGC) and AC coupling may be implemented. This feedback circuit contains two amplifiers  60  and  61  with respective gains of G and A respectively. A low pass filter (integrator)  62  is also added after the feedback amplifier  61 . This type of connection allows the transfer function from input to output to be frequency dependent. By changing the gains of the amplifiers  60  and  61 , the cutoff frequency of this circuit may be varied. The −3 db lower cutoff frequency of the AC coupling is 2piAG. The product of AG must be maintained constant when changing the signal path gain G, in order to keep a constant cutoff frequency. In this manner automatic gain control may be implemented while keeping a constant lower cut-off frequency. As mentioned above the AC coupling provided by this circuit may be effectively changed to DC coupling by making the value of A very small, so that 2piAG is less than 100 HZ. 
   In view of the foregoing, it will be evident to a person skilled in the art, what various modifications may be made in the embodiments given, such as digital signal processing, gain control, channel filtering, and reduction of cut off frequencies. Further the invention is thus not limited to the examples provided.

Technology Classification (CPC): 7