Patent Abstract:
A circuit includes a resonant tunneling device which is responsive to an input signal for causing an electrical signal characteristic to undergo a quantum jump in magnitude that takes an interval of time. According to one feature, a differentiator responds to the quantum jump in magnitude by producing a narrow pulse with a duration which is approximately the interval of time. According to a different feature, a sampling portion responds to the quantum jump in magnitude by sampling a signal during a time period having a duration which is approximately the interval of time.

Full Description:
TECHNICAL FIELD OF THE INVENTION  
       [0001]     This invention relates in general to high-frequency circuits and, more particularly, to high-frequency circuits which utilize a pulse with a very narrow width.  
       BACKGROUND OF THE INVENTION  
       [0002]     In high-frequency circuits, there is often a need for a pulse having a very narrow width. As one example, there are low-noise, phase-locked microwave oscillators which effect phase sampling with a solid-state phase detector. In a known system, the sampling phase detector uses a step recovery diode (SRD) to generate a pulse which has a fairly narrow width, and which is used to clock a diode bridge mixer-phase detector. In particular, the voltage across the SRD is differentiated, in order to generate a pulse that corresponds to a time interval when the SRD voltage has a fairly high slew rate. Although circuits of this type have been generally adequate for their intended purposes, they have not been satisfactory in all respects.  
         [0003]     More specifically, the narrow pulses generated by differentiating an SRD voltage have a width of approximately 22 to 50 picoseconds. While this is sufficiently narrow for many systems, there are other systems which operate at very high frequencies, where even this narrow pulse width is too large, and can produce undesirable effects such as jitter, and/or limits on the gain-bandwidth product.  
       SUMMARY OF THE INVENTION  
       [0004]     From the foregoing, it may be appreciated that a need has arisen for a method and apparatus which avoid at least some of the disadvantages of pre-existing techniques. According to one form of the invention, a method and apparatus are provided to address this need, and involve: providing a circuit having a first portion which includes a resonant tunneling device, and a second portion which includes a differentiator; applying to the first portion an input signal; causing the resonant tunneling device to respond to the input signal by effecting a quantum jump in magnitude of an electrical signal characteristic from a first value to a second value, the second value being substantially different from the first value, and the quantum jump in magnitude from the first value to the second value taking an interval of time; and causing the differentiator to respond to the quantum jump of the electrical signal characteristic from the first value to the second value by producing a narrow pulse having a duration which is approximately equal to the interval of time.  
         [0005]     A different form of the invention involves: providing a circuit having a first portion which includes a resonant tunneling device, and a second portion which includes a sampling portion with a sampling input; applying to the first portion an input signal; applying to the sampling input a signal to be sampled; causing the resonant tunneling device to respond to the input signal by effecting a quantum jump in magnitude of an electrical signal characteristic from a first value to a second value, the second value being substantially different from the first value, and the quantum jump in magnitude from the first value to the second value taking an interval of time; and causing the sampling portion to respond to the quantum jump in magnitude of the electrical signal characteristic from the first value to the second value by sampling the signal at the sampling input during a time period which is approximately equal in duration to the interval of time.  
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0006]     A better understanding of the present invention will be realized from the detailed description which follows, taken in conjunction with the accompanying drawings, in which:  
         [0007]      FIG. 1  is a schematic circuit diagram of an apparatus which is a sampling phase detector circuit that embodies aspects of the present invention;  
         [0008]      FIG. 2  is a graph of a curve that shows how a current flowing through a resonant tunneling diode in the embodiment of  FIG. 1  will vary in response to variation of a voltage applied across it;  
         [0009]      FIG. 3  depicts two related graphs, the upper graph showing how the voltage across the resonant tunneling diode will vary over time as the current through it is progressively increased and then progressively decreased, and the lower graph showing an output voltage that a differentiating portion of the circuit of  FIG. 1  will produce over time in response to the voltage shown in the upper graph;  
         [0010]      FIG. 4  is a schematic circuit diagram of an apparatus which is an alternative embodiment of the apparatus of  FIG. 1 , and which embodies aspects of the present invention;  
         [0011]      FIG. 5  is a schematic circuit diagram of an apparatus which is another alternative embodiment of the apparatus of  FIG. 1 , and which embodies aspects of the present invention;  
         [0012]      FIG. 6  is a graph which depicts a power spectral density in relation to frequency of an output of a resonant tunneling diode in the embodiment of  FIG. 5 ;  
         [0013]      FIG. 7  is a schematic circuit diagram of an apparatus which is still another alternative embodiment of the apparatus of  FIG. 1 , and which embodies aspects of the present invention; and  
         [0014]      FIG. 8  is a schematic circuit diagram of an apparatus which is yet another alternative embodiment of the apparatus of  FIG. 1 , and which embodies aspects of the present invention.  
     
    
     DETAILED DESCRIPTION  
       [0015]      FIG. 1  is a schematic diagram of an apparatus which is a sampling phase detector circuit  10 . The circuit  10  includes an input portion  12 , a differentiating portion  13 , and a sampling portion  14 . The circuit  10  has a reference input defined by a pair of terminals  16  and  17  in the input portion  12 , a sample input defined by a pair of terminals  18  and  19  in the sampling portion  14 , and an output defined by a pair of terminals  21  and  22  in the sampling portion  14 .  
         [0016]     The input portion  12  includes a transformer  26  with an input coil  27  and an output coil  28 . The ends of the input coil  27  are each coupled to a respective one of the input terminals  16  and  17 , and the input terminal  17  is also coupled to ground. The input portion  12  includes a resonant tunneling diode (RTD)  31  of a known type, which is coupled between two nodes  32  and  33  of the circuit. The ends of the output coil  28  of the transformer  26  are each coupled to a respective one of the two nodes  32  and  33 . The input portion  12  also includes a resistor  36  and a capacitor  37 , which are coupled in parallel between the node  32  and ground, and a resistor  38  and a capacitor  39 , which are coupled in parallel between the node  33  and ground. The resistors  36  and  38  are substantially equivalent, and the capacitors  37  and  39  are substantially equivalent.  
         [0017]     The differentiating portion  13  has two capacitors  46  and  47 , which are substantially equivalent, and which effectively serve as a differentiator. The capacitor  46  has one end coupled to the node  32 , and its opposite end coupled to a node  48 . The capacitor  47  has one end coupled to the node  33 , and its opposite end coupled to a node  49 .  
         [0018]     The sampling portion  14  includes two Schottky diodes  51  and  52 , which are equivalent. The diodes  51  and  52  are coupled in series between the nodes  48  and  49 , and a further node  56  is defined between the diodes  51  and  52 . The diodes  51  and  52  are oriented so that the cathode of diode  51  is coupled to the node  48 , and the anode of diode  52  is coupled to the node  49 . The sampling portion  14  has three resistors  61 - 63  which are coupled in series with each other between the nodes  48  and  49 . The resistors  61  and  63  have substantially the same resistance. The resistor  62  is a variable trim resistor, with a slider coupled to the terminal  22  of the output. The resistor  62  can be adjusted so as to maintain balance within the illustrated circuit.  
         [0019]     In the sampling portion  14 , the terminal  18  of the sample input is coupled to ground. A capacitor  71  is coupled between the node  56  and the terminal  19  of the sample input. A resistor  72  is coupled between the node  56  and the terminal  21  of the output, and a capacitor  73  is coupled between the terminal  21  and ground.  
         [0020]     The RTD  31  is a device of a known type, with operational characteristics which are known in the art. Nevertheless, to facilitate an understanding of the present invention, the operational characteristics of the RTD  31  are discussed briefly here.  
         [0021]      FIG. 2  is a graph of a curve that shows how a current flowing through the RTD  31  will vary in response to variation of a voltage applied across the RTD  31 . It will be noted that the current has a resonant peak at  81 , and has a further and larger resonant peak at  82 , which is not visible in its entirety in  FIG. 2 . There is a valley  83  between the two peaks  81  and  82 .  
         [0022]     Although the curve in  FIG. 2  can be viewed as a representation of how current varies as a function of a variation in voltage, it can conversely be viewed as a representation of how voltage varies as a function of a variation in current. In this regard, it will be noted that, as the current through the RTD is progressively increased to a value of I 1  from a value of zero, the voltage progressively increases to a value of V 1  from a value of zero, as indicated diagrammatically at  86 .  
         [0023]     Then, as soon as the current exceeds I 1  the voltage suddenly makes a quantum jump at  87  from a value of V 1  at the top of the resonant peak  81  to a value of V 2  at a point along the leading edge of the resonant peak  82 . As is known in the art, this significant change in voltage from V 1  to V 2  occurs extremely rapidly, for example as fast as 1.5 to 2.0 picoseconds. Then, as the current continues to progressively increase above I 1 , the voltage progressively increases above V 2 , as indicated diagrammatically at  88 .  
         [0024]     Assume that the current is thereafter progressively decreased. The voltage also progressively decreases, as indicated diagrammatically at  91 . The decreasing current eventually reaches a value of I 2 , which corresponds to a voltage V 3 . As soon as the current is decreased below the value I 2 , then the voltage very rapidly makes a quantum jump at  92  from the voltage V 3  to the voltage V 4 , and then continues to progressively decrease, as indicated at  93 . The change at  92  from the voltage V 3  to the voltage V 4  occurs very rapidly, for example in about 1.5 to 2.0 picoseconds. The time intervals of 1.5 to 2.0 picoseconds mentioned above are typical time intervals, but both are determined by the structural configuration of the RTD, and either or both can be varied by adjusting the structural configuration of the RTD.  
         [0025]     The curve shown in  FIG. 2  represents a relationship between a positive current and a positive voltage for the RTD  31 . For a negative current and negative voltage, and as is known in the art, there is a similar curve for the RTD  31 , which is a mirror image of the curve shown in  FIG. 2 , reflected about the origin point at the intersection of the two axes.  
         [0026]     During normal operation, a reference voltage V REF  is applied between the input terminals  16  and  17 . For purposes of the present discussion, this input signal is assumed to be a sine wave, but it could alternatively be some other type of waveform. The transformer  26  responds to this input signal by causing a current to flow through the RTD  31 , where the variation in current flow through the RTD conforms to a sine function.  
         [0027]      FIG. 3  shows two related graphs. The upper graph shows an example of how the voltage across the RTD  31  varies over time, as the current through the RTD  31  is first progressively increased, and then progressively decreased. In this regard, the curve shown in  FIG. 3  has segments  106 - 108  and  111 - 113 , which respectively correspond to  86 - 88  and  91 - 93  in  FIG. 2 . For clarity in the present discussion, the curve segments  106 ,  108 ,  111  and  113  are assumed to correspond to portions of the sine wave where the rate of change is relatively constant, and they are therefore shown in  FIG. 3  as straight lines.  
         [0028]     The curve segment  107  represents the rapid quantum jump in voltage from V 1  to V 2 , and the curve segment  112  represents the rapid quantum drop in voltage from V 3  to V 4 . As discussed above, it is an inherent characteristic of the RTD  31  that the voltage changes at  107  and  112  each occur very rapidly, for example in about 1.5 to 2.0 picoseconds. The voltage across the RTD  31 , such as that shown in the upper graph in  FIG. 3 , serves as the input to the differentiating portion  13  in the circuit of  FIG. 1 , which includes the capacitors  46  and  47 .  
         [0029]     The lower graph in  FIG. 3  shows the output voltage that the differentiating portion  13  will produce over time between the nodes  48  and  49 , in response to the voltage shown in the upper graph in  FIG. 3 . In effect, the curve shown in the lower graph of  FIG. 3  represents the derivative of the curve shown in the upper graph of  FIG. 3 . It will be noted that the rapid voltage change at  107  in the upper graph produces a large positive pulse  121  of very narrow width, and the voltage change at  112  produces a large negative pulse  122  of very narrow width. In the disclosed embodiment, the widths  123  and  124  of the pulses  121  and  122  are each in the range of approximately 1.5 to 2.0 picoseconds, for example about 1.7 picoseconds. Due to the polarity of the diodes  51  and  52 , the diodes recognize one of the pulses  121  and  122  and ignore the other thereof, such that only one of these pulses actually appears at the node  56  which is located between the diodes  51  and  52 .  
         [0030]     A signal which is to be sampled is applied between the terminals  18 - 19  of the sample input, and is referred to here as V SAMPLE . This signal is an alternating current (AC) signal, and is applied to the storage capacitor  73  through the coupling capacitor  71  and the resistor  72 . The voltage across the storage capacitor  73  determines the output voltage V OUT  at the output terminals  21 - 22 . When the node  56  receives a large and narrow pulse from the differentiating portion  13  through the diodes  51  and  52 , the diodes  51  and  52  effectively couple in the load resistors  61 - 63 , SO that a portion of the energy introduced at the sample input  18 - 19  is absorbed in the load resistors  61 - 63 . This deprives the storage capacitor  73  of a portion of the charge that would otherwise end up on the capacitor  73 . Consequently, the pulse from the differentiating portion  13  causes the output voltage V OUT  to be different than it otherwise would have been, which represents a form of sampling of the sample signal V SAMPLE  during the time duration of the narrow pulse received from differentiating portion  13 .  
         [0031]      FIG. 4  is a schematic diagram of an apparatus  140 , which is an alternative embodiment of the apparatus  10  of  FIG. 1 . The apparatus  140  includes an input portion  142  which is different from the input portion  12  of  FIG. 1 , and also includes a differentiating portion  13  and a not-illustrated sampling portion which are respectively identical to the differentiating portion  13  and the sampling portion  14  of  FIG. 1 . In  FIGS. 1 and 4 , equivalent parts are identified with the same reference numerals, and the following discussion addresses the differences between these embodiments.  
         [0032]     The input portion  142  in  FIG. 4  includes the input terminals  16  and  17  of the reference input, and also includes the RTD  31 . The input portion  142  has two terminals  146  and  147 , to which are applied respective direct current (DC) bias voltages +V and −V, which are equal and opposite in magnitude. A field effect transistor (FET)  148  has its source coupled to the terminal  146 , and its drain coupled to one end of a resistor  149 . The other end of the resistor  149  is coupled to the node  32  between the capacitor  46  and the RTD  31 . The gate of the FET  148  is coupled to the node  32 .  
         [0033]     A further field effect transistor (FET)  151  has its source coupled to the node  33  between the capacitor  47  and RTD  31 , and its drain coupled to one end of a resistor  152 . The other end of the resistor  152  is coupled to the terminal  147 . The gate of the FET is coupled to the terminal  16 . In the embodiment of  FIG. 4 , the FETs  148  and  151  are equivalent, and the resistors  149  and  152  have the same resistance. The FET  148  and resistor  149  effectively serve as a current source, and the FET  151  and the resistor  152  effectively serve as a current sink.  
         [0034]     A reference signal is applied to the reference input terminals  16 - 17 , in the form of a voltage which causes dynamic variation in the conductivity of the FET  151 , thereby effecting dynamic variation of the amount of current flowing through the FET  148 , the resistor  149 , the RTD  31 , the FET  151 , and the resistor  152 . Thus, the voltage at the terminals  16 - 17  is effectively converted into a varying current through the RTD  31 , which causes the RTD  31  to produce a voltage between the nodes  32  and  33  which is similar to the voltage shown in the upper graph of  FIG. 3 . The differentiating portion  13  and not-illustrated sampling portion of the embodiment of  FIG. 4  operate the same as their counterparts in the embodiment of  FIG. 1 , and are therefore not described here in detail.  
         [0035]      FIG. 5  is a schematic diagram of an apparatus  160  which is another alternative embodiment of the apparatus  10  of  FIG. 1 . The apparatus  160  includes an input portion  162  which is different from the input portion  12  of  FIG. 1 , and also includes a differentiating portion  13  and a not-illustrated sampling portion which are respectively identical to the differentiating portion  13  and the sampling portion  14  of  FIG. 1 . In  FIGS. 1 and 5 , equivalent parts are identified with the same reference numerals, and the following discussion addresses the differences between these embodiments.  
         [0036]     In the input portion  162  of  FIG. 5 , the node  33  between the RTD  31  and the capacitor  47  is coupled to one end of a resistor  164 , and the other end of the resistor  164  is coupled to ground. A resistor  166  has one end coupled to the node  32  between the capacitor  46  and the RTD  31 , and its other end coupled to a node  167 . The resistors  164  and  166  have the same resistance. The FET  151  has its source coupled to the terminal  146 , its drain coupled to the node  167 , and its gate coupled to the terminal  16 . The terminal  17  is coupled to ground. A further FET  171  has its source coupled to the node  167 , its drain coupled to the terminal  147 , and its gate coupled to its own drain. The FET  171  is equivalent to the FET  151 . The FET  171  serves as a form of constant current source, which operates substantially independently of changes in the voltage applied across it. Since the current flowing through the FET  171  is constant but the current flowing through the FET  151  is not, variation of the current through the FET  151  operates through the resistor  166  to vary the current flowing through the RTD  131 .  
         [0037]     As in the input portions of the other embodiments discussed above, the circuitry of the input portion  162  takes the voltage of the reference signal applied at the terminals  16 - 17  of the reference input, and converts it into a corresponding current flow through the RTD  31 . This causes the RTD  31  to generate between the nodes  32  and  33  a voltage comparable to that shown in the upper graph of  FIG. 3 . The differentiating portion  13  and the not-illustrated sampling portion of the embodiment  160  operate in the same manner as their counterparts in the embodiment of  FIG. 1 , and their operation is therefore not described here in detail.  
         [0038]      FIG. 6  is a graph which depicts an operational characteristic of the circuit of  FIG. 5 . In particular,  FIG. 6  shows the power spectral density of the output of the RTD  151  (vertical axis), in relation to frequency (horizontal axis). This characteristic is determined mathematically by multiplying the Fourier transform of the voltage across the RTD by its complex conjugate. The units along the X-axis represent frequency/200 MHz. The units along the Y-axis are dBc, or in other words Decibels relative to the power in the input carrier to the circuit. The curve of  FIG. 6  corresponds to application of a 10 GHz sine wave to the input of the FET  151 . The two FETs  151  abd  171  serve as a non-inverting buffer of this signal, and the buffered output is applied to the resistor  166 . The resistor  166  converts this voltage into a current, which is used as a sinusodial bias current to the RTD.  
         [0039]     As evident from  FIG. 6 , the effective output of the RTD is rich in harmonics, up to and above 200 GHz. In particular, these harmonics are seen in the plot as strong, discrete peaks in the power spectral density at various frequencies. Peaks are visible at the fundamental frequency (10 GHz), and at even and odd harmonics up to 190 GHz. Actually, the slow drop in spectral power with increasing frequency shows that the RTD waveform provides a very narrow pulse that will approximate an ideal impulse generator, running at the frequency of the input (which in this example case is 10 GHz). The harmonics are desirable for certain applications, for example where a circuit of the type shown in  FIG. 5  is used as part of a low noise, phase-locked microwave oscillator. The harmonics permit phase lock to be accurately and reliably achieved at frequencies which are multiples of the fundamental frequency.  
         [0040]      FIG. 7  is a schematic diagram of an apparatus  180 , which is still another alternative embodiment of the apparatus  10  of  FIG. 1 . The apparatus  180  includes an input portion  182 , which is different from the input portion  12  of  FIG. 1 , and also includes a differentiating portion  13  and a not-illustrated sampling portion, which are respectively identical to the differentiating portion  13  and the sampling portion  14  of  FIG. 1 . IN  FIGS. 1 and 7 , equivalent parts are identified with the same reference numerals, and the following discussion addresses the differences between these embodiments.  
         [0041]     In the input portion  182  of  FIG. 7 , a reference input defined by terminals  186  and  187  is provided in place of the reference input terminals  16 - 17  of  FIG. 1 . The reference input voltage V REF  is applied to the terminal  186 , and its complement is applied to the terminal  187 .  
         [0042]     A resistor  191  has a first end coupled to the node  33  between the capacitor  47  and the RTD  31 , and has its other end coupled to the terminal  187 . An additional RTD  192  has one end coupled to the node  32  between the capacitor  46  and the RTD  31 , and has its other end coupled to one end of a resistor  193 . The other end of the resistor  193  is coupled to the terminal  186 . A reference current source  196  is coupled between the node  32  and ground. The RTDs  31  and  192  are equivalent, and the resistors  191  and  193  are equivalent.  
         [0043]     Like the input portions of the other embodiments described above, the input portion  182  takes the reference input signal and converts it into a corresponding current flow through the RTD  31 , so that the RTD  31  produces between the nodes  32 - 33  a voltage of the type shown in the upper graph of  FIG. 3 . The differentiating portion  13  and the not-illustrated sampling portion of the apparatus  180  operate in the same manner as their counterparts in the apparatus  10  of  FIG. 1 , and their operation is therefore not discussed here in detail.  
         [0044]      FIG. 8  is a schematic diagram of an apparatus  210 , which is an alternative embodiment of the apparatus  10  of  FIG. 1 . The apparatus  210  includes an input portion  212  which is different from the input portion  12  of  FIG. 1 , and also includes a differentiating portion  13  and a not-illustrated sampling portion which are respectively identical to the differentiating portion  13  and the sampling portion  14  of  FIG. 1 . In  FIGS. 1 and 8 , equivalent parts are identified with the same reference numerals, and the following discussion addresses the differences between these embodiments.  
         [0045]     In the input portion  212 , the input terminals  16 - 17  and the transformer  26  of  FIG. 1  have been replaced with a photodiode  216  and a light source  218 . The photodiode  216  is a component of a known type, such as a PIN photodiode or a metal-semiconductor-metal (MSM) photodiode. The photodiode has its anode coupled to the node  32 , and its cathode coupled to the node  33 . The light source  218  is a periodic pulsed laser of a type known in the art, such as a mode-locked laser, or a fiber-ring laser. Alternatively, the light source  218  could be a continuous laser with a mechanical shutter, or some other device that produces a periodic optical signal. The light source  12  outputs a varying optical signal  221 , which serves as a clock signal that varies in a periodic manner. The optical clock signal  221  causes the photodiode  216  to alternate between conducting and non-conducting states. When the photodiode is in its conducting state, it effectively creates an electrical short across the RTD  31 , so that the voltage across the RTD  31  is very low or zero. When the photodiode switches to its non-conducting state, current from the bias arrangement will cause a current to develop throught the RTD  31 , and the voltage across the RTD  31  will under a quantum jump such as that shown at  87  in  FIG. 2 . In other respects, the operation of the circuit of  FIG. 8  is generally similar to the operation of the circuit  10  of  FIG. 1 , and is therefore not described here in further detail.  
         [0046]     The present invention provides a number of advantages. One such advantage results from the generation of a pulse of very narrow width through use of a resonant tuning diode with a high slew rate, where the slew rate is on the order of about 3 picoseconds per volt. This is five to ten times faster than the slew rate of the step recovery diodes (SRDs) used in pre-existing systems. Therefore, when the voltage across the RTD is differentiated, the result is a pulse with a very narrow width, which can be as much as {fraction (1/35)} of the width of the typical pulse produced in pre-existing systems using SRDs. The ability to generate a very narrow pulse is advantageous in a variety of applications. As one example, when used in the context of a very fast sampling phase detector for a low-noise phase-locked microwave oscillator, the narrow pulse provides more accurate sampling, along with a reduction in jitter and an increase in bandwidth, where the bandwidth can be as much as five to ten times better than in comparable pre-existing systems which utilize SRDs. By using an RTD to generate a narrow pulse, sampling can occur at frequencies of 100 GHz to 200 Ghz, which was not possible with the wider pulses generated in pre-existing systems using SRDs.  
         [0047]     Although several selected embodiments have been illustrated and described in detail, it will be understood that various substitutions and alterations can be made without departing from the scope of the present invention. That is, the depicted circuits are merely exemplary, and it is possible to add, delete, and/or rearrange components, or to utilize different circuit configurations, while still realizing the present invention. Other substitutions and alterations are also possible without departing from the spirit and scope of the present invention, as defined by the following claims.

Technology Classification (CPC): 1