Patent Abstract:
An electrosurgical generator for supplying RF power to an electrosurgical instrument for cutting or vaporising tissue has an RF output stage ( 42 ) with an RF power bridge (Q 1 , Q 2 , Q 3 , Q 4 ), a pair of output lines ( 74 ) and a series-resonant output network ( 48 ). The output impedance of the output stage ( 42 ) at the output lines ( 74 ) is less than 200/√P ohms, where P is the maximum continuous RF output power of the generator. The generator offers improved cutting and vaporising performance, especially in relation to the reliability with which an arc can be struck when presented with an initial low impedance load. Overloading of the output stage is prevented by rapidly operating protection circuitry responsive to a predetermined electrical condition such as a substantial short-circuit across the output lines. In the preferred embodiment, the output stage is capable of maintaining output pulses at least 1kW peak by supplying the power bridge from a large reservoir capacitor ( 60 ). Pulsing is dynamically variable in response to load conditions by controlling the maximum energy per pulse in response to the reservoir capacitor voltage.

Full Description:
This nonprovisional application claims the benefit of U.S. Provisional Application No. 60/449,859, filed Feb. 27, 2003 and No. 60/505,756 filed Dec. 3, 2003. 

   FIELD OF THE INVENTION 
   This invention relates to an electrosurgical generator for supplying radio frequency (RF) power to an electrosurgical instrument, and primarily to a generator having a series-resonant output network. 
   BACKGROUND OF THE INVENTION 
   Conventionally, electrosurgical generators make use of a configuration comprising a voltage source coupled to an electrosurgical instrument via a coupling capacitor which defines a matched output impedance between 50 and 500 ohms. Such a configuration produces a power-versus-load impedance characteristic having a power maximum at a matched impedance, with power falling off progressively on each side of this peak. In practice, when conducting electrosurgery, the load impedance can change over a very wide range, resulting in unpredictable clinical effects. 
   To deal with this problem, it is known to provide an RF output stage capable of providing an impedance match over a wide range. This has the disadvantage that rapid load impedance changes can produce large output voltage excursions. An alternative approach is to control the DC supply to the RF output stage in response to feedback signals in order that the delivered power is virtually continuous. This may be done by adjusting the power supply DC voltage or by maintaining the supplied DC power constant. These techniques lead to a power versus load impedance characteristic which is virtually flat over a range of impedances, but one limitation is that it is difficult to control the delivery of energy when initiating tissue cutting or vaporisation (as opposed to tissue coagulation). To cut or vaporise tissue using radio frequency power, the initial low impedance load presented by the tissue or surrounding fluid needs to be brought to a higher impedance in order to strike an arc. Delivering too much energy can result in bums adjacent the operative site, excessive smoke, or instrument failure. Delivering too little energy causes a significant delay and can result in unwanted tissue coagulation. 
   It is also known to use an electrosurgical generator to supply a bipolar electrosurgical instrument with pulsed electrosurgical power at very high voltages, e.g. in the region of 1 kilovolt peak-to-peak when removing tissue at an operation site immersed in a conductive liquid, such as saline. The instrument may have an active electrode located at its extreme end to be brought adjacent to or into contact with tissue to be treated, and a return electrode set back from the active electrode and having a fluid contact surface for making an electrical connection with the conductive liquid. To achieve tissue removal, the conductive liquid surrounding the active electrode is vaporised to cause arcing at the electrode. The high voltages used to achieve tissue cutting or vaporisation under varying load impedance conditions are particularly demanding of the generator when the instrument experiences a low load impedance. Indeed, as stated above, under such conditions it is difficult reliably to initiate arcing without unwanted effects. Steps have been taken to increase power density at the active electrode and, hence, improve the reliability with which arcing is started, by reducing the size of the electrode and by roughening its surface, e.g. by applying an oxide layer. The latter technique has the effect of trapping vapour in the irregularities in the surface as a means of increasing power density. 
   It has been found that operation of such instruments at high voltages tends to cause erosion of the active electrode. The rate of erosion increases as the supply voltage is increased, and is also exacerbated by reducing the size of the electrode and providing a roughened surface, as just mentioned. 
   Published European Patent Application No. EP1053720A1 discloses a generator for generating high electrosurgical voltages. 
   SUMMARY OF THE INVENTION 
   According to a first aspect of the present invention, an electrosurgical generator for supplying RF power to an electrosurgical instrument comprises an RF output stage having a least one RF power device, at least one pair of output lines for delivering RF power to the instrument, and a series-resonant output network coupled between the RF power device and the said pair of output lines, wherein the output impedance of the output stage at the output lines is less than 200/√P ohms, where P is the maximum continuous RF output power of the generator in watts. When the generator is configured for wet field surgery, e.g. for use with the electrode or electrodes of the instrument immersed in a conductive fluid such as saline, the maximum continuous power is preferably in the region of 300 W to 400 W. Accordingly, if the maximum output power is 400 W, the output impedance is less than 10 ohms. Dry field electrosurgery, i.e. with the electrode or electrodes not normally immersed, requires less RF output power. In this case, the generator may be configured such that the maximum continuous RF output power is in the region of 16 W, in which case that the output impedance is then less than 50 ohms. In both such cases, the figures are obtained when operating with an output voltage for cutting or vaporising tissue, i.e. at least 300V peak. The output impedance is preferably less than 100/√P ohms, which yields maximum output impedance values of 5 ohms and 25 ohms at the above power outputs. 
   It will be understood that when the RF output of the generator is pulsed, i.e. when RF energy is supplied to a load in bursts, generally as an RF sine wave, the maximum continuous power is the average power measured over several such bursts. 
   According to another aspect of the invention, an electrosurgical generator for supplying RF power to an electrosurgical instrument for cutting or vaporising tissue comprises an RF output stage having: at least one RF power device, at least one pair of output lines for delivering RF power to the instrument, and a series-resonant output network coupled between the RF power device and the said pair of output lines, the network being configured such that the maximum rate of rise of the output current at the output lines is less than (√P)/4 amps per microsecond, P being as defined above. 
   Accordingly, for a typical maximum continuous RF output power of 400 W for wet field electrosurgery, the maximum rate of rise of the output current amplitude, generally when the output lines are short-circuited at the maximum power setting of the generator, is less than 5 A/μs. With P at a typical value of 16 W for dry field electrosurgery, the rate of rise of the output current amplitude is less than 1 A/μs. 
   In a preferred generator in accordance with the invention, there is protection circuitry responsive to a predetermined electrical condition indicative of an output current overload, e.g. due to short-circuiting of the output lines, substantially to interrupt the RF power supplied to the series-resonant output network. The protection circuitry is responsive to short-circuiting with sufficient speed that the supply of RF power to the output network is cut off within a time period corresponding to no more than 20 cycles of the delivered RF power. The protection circuitry is preferably much faster, e.g. being operable to interrupt power delivery within 3 cycles or even 1 cycle. The effect of the series-resonant output network is to delay the build up of current in a fault condition such as when a very low impedance or a short circuit appears across the output lines. The applicants have found that an impedance transition from open to short circuit results in an effective short circuit across the RF power device only after several RF cycles. By arranging for the protection circuitry to respond quickly, the output stage can be disabled before that happens. In general, the protection circuitry is responsive to application of a short-circuit across the output lines sufficiently quickly to disable the RF power device before the current passing therethrough rises to a rated maximum current as a result of the short-circuit. 
   The use of an RF output stage with a relatively low output impedance means that the RF voltage output is substantially directly related to the DC supply voltage applied to the output stage (specifically to the RF power device or devices which it contains). In the preferred embodiment of the invention, each RF power device is operated in a switching mode with the result that a square wave output is applied to the series-resonant output network. The RMS voltage available at the output lines is directly proportional to the supply voltage. It follows that the maximum peak-to-peak output voltage is determined by the DC supply voltage and dynamic feedback to control output voltage is, as a result, not required in this embodiment. 
   The protection circuitry is preferably capable of disabling the output stage within one-and-a-half RF periods after onset of the above-mentioned predetermined electrical condition. Preferably, the predetermined electrical condition is indicative of an instantaneous current in the output stage exceeding a predetermined level, and the speed of response of the protection circuitry is such that the breaching of the predetermined level by the instantaneous current is detected during the same RF cycle that it occurs. Such detection may be performed by current sensing circuitry including a pick-up arrangement, which is typically a current transformer, coupled in series between the RF power device or devices and the series-resonant output network, and a comparator having a first input coupled to the pick-up arrangement (e.g. to the secondary winding of the transformer) and a second input coupled to a reference level source. The reference level source may be a voltage representation of the instantaneous current, i.e. substantially without filtering, in order to cause a change of state of the comparator output within the same RF half-cycle that the threshold is first exceeded, or within the subsequent half-cycle, depending on whether or not full wave rectification is applied ahead of the comparator. The predetermined instantaneous output level is preferably at least 5 A for wet field electrosurgery, and typically 15 A. The output of the comparator is coupled to disabling circuitry to disable the power device or devices when the comparator output changes state in response to the instantaneous current sensed by the pick-up arrangement exceeding the predetermined level as set by the reference source. The current shut-down aspect of the protection circuitry is not limited by impedance. 
   Generally, it is necessary only to interrupt power delivery for a short time. Consequently the protection circuitry includes a monostable stage and is operable in response to detection of the predetermined condition to disable the power device for a limited period determined by a time constant of the monostable stage which is typically less than 20 cycles of the operating frequency of the generator. 
   Preferably, the generator has an RF source coupled to the power device or devices, the source including an oscillator defining the operating frequency of the generator. The series resonant output network is tuned to this operating frequency. Generally, the source is arranged such that the operating frequency is substantially constant (e.g. during any given treatment cycle). 
   The preferred generator is arranged such that, for a given user setting, the RMS RF output voltage is substantially within a load impedance range of from 600/√P ohms to 1000 ohms, where P is as defined above. Thus, for instance, the RMS RF output voltage constant during each burst of RF energy is maintained to within 20 percent of a maximum value. This can be achieved partly as a result of the series-resonant configuration of the output network. 
   To maintain the constant peak output voltage at low impedances, according to a particular preferred feature of the invention, the RF power supply to the output stage includes a charge-storing element, preferably a capacitance in excess of 1 mF, the output devices being pulsed by a pulsing circuit so that they supply RF energy in bursts with the timing of the bursts, particularly the termination of each burst, being controlled in response to the output of a voltage sensing circuit coupled to the capacitance. The DC power supply voltage to the output stage is preferably 100V or greater. To avoid substantial decay of the supply voltage, the voltage sensing and pulsing circuits are arranged to terminate the individual pulses of RF energy when the sensed voltage falls below a predetermined level, typically set such that pulse termination occurs when the voltage falls by a predetermined percentage value of between 5 percent and 20 percent which, typically, corresponds to the peak RF voltage delivered at the output lines falling to a value between 25V and 100V below its starting value for the respective pulse. The RF energy delivered during each pulse is typically 60 joules for wet field electrosurgery and 2 joules for dry field electrosurgery. Peak power typically reaches at least 1 kW, and preferably 4 kW. 
   The very high peak power capability of the preferred wet field generator (in excess of 1 kW) allows the impedance transition occurring at the start of a tissue cutting or vaporisation cycle to be completed very quickly since only voltages in excess of those required for arcing are delivered. This significantly reduces the delay and the unwanted coagulation effects of some prior art generators. The substantially constant voltage delivery leads to cutting or vaporisation occurring at consistent rates, regardless of changes in tissue type or engagement. 
   According to a further aspect of the invention, there is provided an electrosurgical generator for supplying radio frequency (RF) power to an electrosurgical instrument, wherein the generator comprises an RF output stage having at least one RF power device, at least one pair of output lines for delivering RF power to the instrument, and a series-resonant output network coupled between the RF power device and the output lines, the generator further comprising protection circuitry responsive to a short circuit across the output lines, wherein the output impedance of the output stage is less than 200/√P ohms, where P is the maximum continuous RF output power of the generator in watts, and wherein the protection circuitry is responsive to the said short circuit sufficiently quickly to disable the power device before the current passing therethrough rises to a rated maximum current as a result of the short circuit. The or each power device may be disabled in response to application of the short-circuit to the output lines in a time period corresponding to less than three RF cycles. 
   Another aspect of the invention provides an electrosurgical generator for supplying radio frequency (RF) power to an electrosurgical instrument for cutting or vaporising tissue in wet field electrosurgery, wherein the generator comprises an RF output stage having: at least one RF power device, at least one pair of output lines for delivering RF power to the instrument, and a series-resonant output network coupled between the RF power device and the said pair of output lines, and wherein the output impedance of the output stage at the output lines is less than 10 ohms. 
   Yet another aspect of the invention provides an electrosurgical generator for supplying radio frequency (RF) power to an electrosurgical instrument for cutting or vaporising tissue in dry field electrosurgery, wherein the generator comprises an RF output stage having: at least one RF power device, at least one pair of output lines for delivering RF power to the instrument, and a series-resonant output network coupled between the RF power device and the said pair of output lines, and wherein the output impedance of the output stage at the output lines is less than 50 ohms. 
   According to a yet further aspect of the invention a generator for supplying RF power to an electrosurgical instrument for cutting or vaporising tissue comprises an RF output stage having: at least one RF power device, at least one pair of output lines for delivering RF power to the instrument, and a series-resonant output network coupled between the RF power device and the said pair of output lines, wherein the generator is configured to be capable of maintaining a peak output voltage of at least 300V over a load impedance range of from 600/√P ohms to 1000 ohms, where P is the rated output power in watts. The rated output power is as defined in the International Electrotechnical Commission standard, IEC 60601-2-2. 
   According to yet a further aspect of the invention, there is provided an electrosurgical generator for supplying RF power to an electrosurgical instrument for cutting or vaporising tissue, wherein the generator comprises an RF output stage having: at least one RF power device, at least one pair of output lines for delivering RF power to the instrument, and a series-resonant output network coupled between the RF power device and the said pair of output lines, wherein the generator further comprises a power supply stage coupled to the RF output stage, the power supply stage having an energy storage capacitor capable of storing between 3 percent and 30 percent of the maximum continuous power P (in watts) of the generator in joules. 
   In another aspect of the invention, the energy delivery per pulse (in joules) is between 1 percent and 10 percent of the maximum continuous RF output power (in watts). 
   The invention also includes an electrosurgical generator for supplying RF power to an electrosurgical instrument for cutting or vaporising tissue, wherein the generator comprises an RF output stage having: at least one RF power device, at least one pair of output lines for delivering RF power to the instrument, and a series-resonant output network coupled between the RF power device and the output lines, and wherein the generator further comprises a pulsing circuit coupled to the output stage for pulsing the delivered RF power in such a way that the crest factor of the voltage developed across the output lines increases as the load impedance presented to the output lines decreases whilst the peak output voltage during pulses is maintained at a value greater than 300V. For wet field electrosurgery, the output impedance of the output stage is preferably less than 10 ohms and the crest factor varies by a ratio of at least 2:1 over a load impedance range of from 600/√P to 1000 ohms (typically from 10 ohms to 1000 ohms). For dry field electrosurgery, the output impedance figure is less than 50 ohms, and the crest factor varies by a ratio of at least 2:1 over a load impedance range of 600/√P to 50 kilohms (typically from 50 ohms to 50 kilohms). 
   By “crest factor” we mean the ratio of the peak voltage to the RMS voltage. In the case of a pulsed output waveform, the measurement is conducted over plurality of pulses. 
   According to a tenth aspect of this invention, an electrosurgical generator comprises a source of radio frequency (RF) energy, an active output terminal, a return output terminal, a DC isolation capacitance between the source and the active output terminal, and a pulsing circuit for the source, wherein the source and the pulsing circuit are arranged to generate a pulsed RF output signal at the output terminals, which signal has a peak current of at least 1 A, a simultaneous peak voltage of at least 300V, a modulation rate of between 5 Hz and 2 kHz, and a pulse length of between 100 μs and 5 ms. In preferred embodiments of the invention, the signal has a peak current of at least 3 A. 
   With such a generator it is possible to start arcing even under conditions of relatively low load impedance. Once an arc is established, the load impedance tends to rise, to the extent that the arcing can be maintained using a continuous RF output waveform. Improved power density is available at the active electrode for vaporisation, whilst reducing electrode erosion. 
   The length of the pulses is preferably between 0.5 ms and 5 ms, the pulse duty cycle typically being between 1% and 20% and, more preferably, between 2% and 10%. 
   The preferred generator in accordance with the invention has a resonant output network and is operable to generate, e.g. during at least an initial part of a treatment period, a peak power of at least one kilowatt, and typically at least 3 or 4 kilowatts. Improvements in electrode erosion performance can be achieved by providing means in the generator for limiting the output voltage to a value in the region of 900V to 1100V peak-to-peak. 
   In the preferred generator, the source and the pulsing circuit are arranged to generate, in an initial period, a pulsed RF output signal at the output terminals, which signal has a peak current of at least 1 A, a simultaneous peak voltage of at least 300V, a modulation rate of between 5 Hz and 2 kHz, and a pulse length of between 100 μs and 5 ms, and, in a subsequent period, to generate a constant power RF output signal at the output terminals. 
   Different ways of causing the generator to end the above-mentioned initial period of operation and begin the so-called subsequent period are feasible. One generator embodiment is arranged such that the switchover from the initial period to the subsequent period occurs automatically at a predetermined time interval after the beginning of the initial period. In an alternative embodiment, the generator has means for monitoring, in use of the generator, the load impedance between the active and return output terminals, and is arranged to cause switchover to the subsequent period when the magnitude of the output impedance increases by a predetermined factor, typically between 5 and 20, and preferably 10, or when it exceeds a predefined threshold. 
   The preferred generator uses a third switching-over technique involving the charge-storing element mentioned above. In this case, the source of RF energy includes an RF output stage, and the generator has a power supply including the charge-storing element such as a large capacitor for supplying power to the output stage. When the treatment period includes an initial period and a subsequent period, as described above, the capacitor is used to supply power at least during the initial period. Associated with the charge-storing element is a voltage-sensing circuit for sensing the voltage supplied to the output stage by the charge-storing element, the generator being arranged such that treatment ends or the subsequent period begins in response to the supply voltage as sensed by the voltage-sensing circuit reaching a predetermined voltage threshold. Indeed, it is possible to control the length and timing of individual pulses using the same voltage-sensing circuit. In this case, the voltage-sensing circuit forms part of the above-mentioned pulsing circuit and the timing of at least the beginnings of the pulses produced by the output stage during the initial period being determined in response to the supply voltage reaching the above-mentioned voltage threshold. It is possible to arrange for both the leading and trailing edges of the pulses produced by the output stage to be determined by the supply voltage respectively falling below and exceeding the respective voltage thresholds. 
   The charge-storing capacitance is preferably at least 1000 μF and advantageously has a capacity in excess of 5 J. 
   As already stated, the preferred generator has a tuned output. Indeed, good results have been obtained using a generator with a resonant output network, the load curve of the generator (i.e. the curve plotting delivered power versus load impedance) having a peak at a load impedance below 50 ohms. Delivery of peak power levels into low load impedances is aided by forming the output network as a series-resonant network comprising the series combination of an inductance and a capacitance, the output of the network being taken across the capacitance. The output may be taken to all output terminal of the generator via a coupling capacitor and, optionally, a step-up transformer from a node between the inductance and the capacitance of the series combination. Whilst it is possible, instead, to take the output from across the inductance, taking it across the capacitor has the advantage of reducing switching transients. As a further alternative, the generator may have its output terminals connected to the resonant output network so that, effectively, when a load is connected to the terminals it is connected as an impedance in series with the inductance and capacitance forming the resonant combination, e.g. between the inductance and the capacitance. 
   The resonant output network typically provides a source impedance at the output terminals in the range of from 50 ohms to 500 ohms. 
   Not least because the resonant frequency of the output network can vary with load impedance as a result of coupling capacitance, the RF source may include a variable frequency RF oscillator, the output frequency advantageously being limited to a maximum value below the resonant frequency of the output network when connected to a matching load impedance, i.e. a load impedance equal to its source impedance. 
   The generator may be combined with a bipolar electrosurgical instrument to form an electrosurgical system, the instrument having at least an active electrode coupled to the active output terminal of the generator and a return electrode coupled to a generator return output terminal. The invention has particular application to an electrosurgery system in which the bipolar electrosurgical instrument has an active electrode formed as a conductive, preferably U-shaped loop. Such a loop is often used for excising tissue samples but places particular demands on the generator in terms of achieving saline vaporisation and arcing. 
   According to an eleventh aspect of the invention, an electrosurgery system comprises a generator having a source of radio frequency (RF) energy and, coupled to the generator, a bipolar electrosurgical instrument having an electrode assembly with at least a pair of electrodes, wherein the generator is adapted to deliver RF energy to the electrode assembly in an initial period as a pulse modulated RF signal which, in use with the pair of electrodes, has a peak current of at least 1 A, a simultaneous peak voltage of at least 300V, a modulation rate of between 5 Hz and 2 kHz, and a pulse length of between 100 μs and 5 ms. 
   Again, the system may be adapted to deliver RF energy to the electrode assembly, in an initial period, as a pulse modulated RF signal which, in use with the pair of electrodes, has a peak current of at least 1 A, a simultaneous peak voltage of at least 300V, a modulation rate of between 5 Hz and 2 kHz, and a pulse length of between 100 μs and 5 ms, and to deliver RF energy to the electrode assembly in a subsequent period as a continuous power RF signal. The peak current is preferably at least 3 A. 
   The invention will be described below by way of example with reference to the drawings. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     In the drawings: 
       FIG. 1  is a general diagram showing an electrosurgery system including a generator in accordance with the invention and a bipolar electrosurgical instrument; 
       FIGS. 2A and 2B  are respectively perspective and side views of a loop electrode assembly forming part of the bipolar instrument shown in  FIG. 1 ; 
       FIG. 3  is a block diagram illustrating the main components of the generator; 
       FIG. 4  is a simplified circuit diagram of an RF output stage forming part of the generator; 
       FIG. 5  is an illustrative load curve for the generator of  FIG. 1 ; 
       FIG. 6  is a more detailed circuit diagram of the RF output stage; 
       FIG. 7  is a block diagram of an alternative electrosurgical generator in accordance with the invention; 
       FIG. 8  is a circuit diagram of a resonant output network of the alternative generator; and 
       FIG. 9  is the load curve of the generator of  FIG. 7 . 
   

   DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
   Referring to  FIG. 1 , a generator  10  has an output socket  10 S providing a radio frequency (RF) output for an electrosurgical instrument in the form of an endoscope attachment  12  via a connection cord  14 . Activation of the generator may be performed from the instrument  12  via a control connection in cord  14  or by means of a footswitch unit  16 , as shown, connected separately to the rear of the generator  10  by a footswitch connection cord  18 . In the illustrated embodiment, the footswitch unit  16  has two footswitches  16 A and  18 B for selecting a coagulation mode and a cutting mode of the generator respectively. The generator front panel has push buttons  20  and  22  for respectively setting coagulation and cutting power levels, which are indicated in a display  24 . Push buttons  26  are provided as alternative means for selection between coagulation and cutting modes. The instrument  12  has a detachable loop electrode assembly  28  with a dual electrode structure and is intended for use in a saline field. 
   The instrument  12  has a detachable loop electrode assembly  28  with a dual electrode structure and intended for use in a saline field.  FIGS. 2A and 2B  are enlarged views of the distal end of the electrode assembly  28 . At its extreme distal end the assembly has a U-shaped loop electrode  30  depending from a pair of electrode assembly arms  32  which are mounted side-by-side in a clip  34  intended to be snapped onto an endoscope. The loop electrode  30  is an active electrode. Each of the arms  32  is formed as a coaxial cable, the exposed conductive outer shield of which, in each case, forms a return electrode  36 . In operation immersed in a saline field, the loop electrode  30  is typically used for excising tissue samples, the electrosurgical voltage developed between the loop electrode  12 A and fluid contacting surfaces of the return electrodes  36  promoting vaporisation of the surrounding saline liquid at the loop electrode  30 , and arcing through the vapour envelope so formed. 
   The loop electrode  30  comprises a composite molybdenum rhenium wire with an oxide coating to promote increased impedance in the electrode/fluid interface and, as a result, to increase power density at the surface of the electrode. 
   The width of the loop is typically in the range of 2.5 mm to 4 mm and the wire typically has a diameter in the range of 0.20 to 0.35 mm. 
   This loop electrode assembly places particular demands on the generator in terms of starting vaporisation and arc formation. 
   Efforts to improve the starting of the arc (the “firing up”) of this electrode assembly by reducing the wire diameter and forming oxide layers have tended to increase the rate of erosion or resulted in the loop being mechanically flimsy. 
   It should be noted that generators in accordance with the invention not limited to use with a loop electrode assembly, nor to use in wet field surgery. 
   The generator will now be described in more detail with reference to  FIG. 3 . It has an RF source in the form of an oscillator  40  which is connectible to an RF output stage  42 . The output stage  42  comprises a mosfet power bridge forming part of a power mosfet and driver circuit  44 , a current sensing element  46  and a resonant output network  48 . The oscillator  40  is configured to operate at a substantially constant RF frequency and the output network  48  is tuned to that frequency. In general terms, the RF source coupled to the RF power device or devices defines the operating frequency of the generator, and the output network (which, as will be described below, is series-resonant) is tuned to the operating frequency. In this embodiment of the invention the operating frequency is substantially constant. 
   Power to the RF output stage  42 , or, more specifically, to the power mosfets, is supplied from a DC power supply  50  via a supply rail  58 . A 4.7 mF reservoir capacitor  60  is connected between the supply rail  58  and ground. The voltage on the supply rail  58  is sensed by a voltage sensing circuit  62  which controls a first transmission gate  64  connected in series between the RF oscillator  40  and driver devices in the power mosfet and driver circuit  44 . 
   The current sensing element  46  in the output stage  42  is a series-connected current transformer, the secondary winding of which is coupled to a first input of a comparator  66  which also receives on the other of its inputs a reference signal from a reference input  68 . The output of the comparator controls a monostable  70  which, in turn, controls a second transmission gate  72  coupled in series with the gate  64  in the path between the oscillator  40  and the drivers in the power mosfet and driver circuit  44 . The output network  48  supplies RF power to an output termination  74  which, in practice, is a pair of output lines, as will be described hereinafter. Operation of the generator is normally pulsed insofar as RF energy is supplied to the output lines  74  in bursts controlled by the combination of the voltage sensing circuit  62  and gate  64  which operates as part of a pulsing circuit. When the generator is activated, the voltage on the supply rail  58  tends to fall, at least when the load impedance coupled across output lines  74  is relatively low, owing to the discharge of reservoir capacitor  60 . When the DC supply voltage on the supply rail  58  falls to a preset value, the output of the voltage sensing circuit  62  changes state and transmission gate  64  is driven to its open circuit condition, thereby disabling the power mosfet and driver circuit  44 . The reservoir capacitor  60  then recharges and the voltage sensing circuit  62  causes the gate  64  to reconnect the oscillator  40  when the supply rail voltage reaches a second, higher present value. In this way it is possible to control the amount of energy delivered in each pulse. 
   The current sensing element  46 , the comparator  66 , the monostable  70  and the second transmission gate  72  act together as a protection circuit to protect the mosfet power devices in circuit  44  against excessive current drain caused, for instance, by a short circuit across the output lines  74 . The storage of energy in output network  48  delays the transfer of the short circuit across the output lines  74  to the power devices in the mosfet and driver circuit  44 . 
   The electrical circuit condition sensed by the current sensing element  46  and the comparator  66  is the current flowing between the power mosfets in circuit  44  and the output network  48  rising to a level which could be indicative of a short circuit having been applied across the output lines  74 . When the current reaches a preset current level, as detected by the comparator  66 , the comparator output changes state and the monostable  70  causes the second transmission gate  72  to become open circuit, disabling the power mosfets and driver stage  44 . The monostable time constant is typically set to 0.5 seconds or less, which allows a warning signal to be generated for alerting the user. However, owing to energy storage in the series-resonant circuit, it is possible to protect the RF power devices with a monostable time constant of about 20 RF cycles at an operating frequency of 400 kHz. 
   The configuration of the output stage  42  is shown in principle in the simplified circuit diagram of  FIG. 4 . Referring to  FIG. 4 , the power mosfet and driver stage  44  shown in  FIG. 3  has a power mosfet bridge comprising a first push-pull pair of FET power devices Q 1 , Q 2  and a second power FET device push-pull pair Q 3 , Q 4 , each pair having a respective output node which, when the pairs are driven 180° out of phase, produces a square wave at the frequency of the oscillator  40  ( FIG. 2 ) at the input to the series resonant output network  48 . Each pair of power mosfets Q 1 , Q 2 ; Q 3 , Q 4  is coupled between the supply rail  58  and ground. Accordingly, since each of the mosfets is a virtual short circuit when driven “on”, the voltage applied to the output network  48  swings virtually between ground and the supply rail voltages. The reservoir capacitor  60  shown in  FIG. 3  is, of course, connected across the respective power mosfet pairs, as shown in  FIG. 4 . 
   The output network is series-resonant in that an inductor L 1  and a resonating capacitor C 1  are coupled in series between the output nodes  76 ,  78  of the first and second power mosfet pairs respectively. In this embodiment, the load resistance R L  constituted in practice by an electrosurgical instrument coupled between the output lines  74 , and the tissue and/or fluid present across its electrode assembly, is connected in series between inductor L 1  and capacitor C 1 . As explained above, the series-resonant tuned circuit formed by inductor L 1  and capacitor C 1  acts as an energy storing device which delays the current build-up between the nodes of the power mosfet bridge Q 1 –Q 4  should the load resistance R L  drop to a very low value. Another feature of this resonant arrangement is that it is a low impedance at one frequency only, which means that the delivered output signal consists almost exclusively of the fundamental component of the waveform produced by the power mosfets, conditional, of course, upon the frequency of resonance of the network  48  being the same as that of the operating frequency of the oscillator stage  40  ( FIG. 3 ). 
   One of the characteristics given to the generator by the output configuration described above with reference to  FIG. 4  is that, during each burst or pulse of RF energy it has an approximately constant voltage load curve, as shown by the power-versus-load impedance load curve shown in  FIG. 5 . This characteristic is particularly suitable for cutting or vaporisation of tissue since it provides the high power required at low impedance without voltage overshoot. The low output impedance and high current required are provided by the direct coupling of the power mosfets to the supply rail and ground, and by the reservoir capacitor  60 , even if a step-up transformer is coupled between the series-resonant elements L 1 , C 1  and the output lines  74 . It is possible, using this configuration, to keep the output impedance of the generator at the output lines  74  to 2 ohms or less. The implications which this has for peak current delivery in a fault condition leads to the need for a protection circuit such as that referred to above. 
   The RF output stage  42  is shown in more detail in  FIG. 6 . As shown in  FIG. 6 , the current sensing element  46  is a current transformer, coupled in series between one of the output nodes  76 ,  78  of the power mosfet bridge and one of the components L 1 , C 1  of the series resonant output network, in this case between node  76  and the inductor L 1 . In this preferred generator, the normal DC supply voltage on supply rail  58  is about 120V. To strike an arc for the purpose of performing tissue cutting or vaporisation, a peak voltage in excess of 380V may be required. Accordingly, and for isolation purposes, the RF output network  48  includes a step-up isolating transformer TR 1  to lift the peak output voltage to the region of 500V peak. The primary winding of the transformer TR 1  has a tuning capacitor C 2  coupled across it to yield a parallel-resonant circuit tuned to the operating frequency to act as a shunt-connected trap. This improves the rejection of harmonics in the power signal supplied to the output lines  74 ., particularly when the output impedance is high, with the consequent benefit of reduced RFI (RF interference). 
   DC blocking is provided by a coupling capacitor C 3  between the transformer TR 1  secondary winding and one of the output lines  74 . 
   The actual resonant frequency of the output network  48  is the result of several elements, these being (1) the main tuning elements represented by the lumped inductance L 1  and the tuning capacitor C 1 , (2) the transformer leakage inductance and cross-coupling capacitance, (3) the DC blocking capacitance, C 3 , and (4) the inductive and capacitive loading of the connecting cable (not shown) between the output lines  74  and the electrosurgical instrument itself. The delay in the current build-up in a fault condition is due to the energy levels in this tuned arrangement. At resonance, this arrangement has a finite loss that may be represented by a series resistance, albeit a very small one. Dynamically, however, the energy levels in the resonant output network cannot be changed instantly. An impedance transition from an open to short circuit only presents a short circuit to the switching stage after several RF cycles at the operating frequency. The comparator  66  shown in  FIG. 3  is capable of detecting such an impedance transition within 1 to 1.5 cycles of the transition beginning at the output lines  74 . This rapid response, as well as allowing the power mosfet and driver circuit  44  to be shut down before damage occurs, has the effect that the amount of energy delivered during a short circuit fault is very small. 
   Referring again to  FIG. 3  and, in particular, the voltage sensing and output stage pulsing circuits  62 ,  64 , the very high peak powers which are achievable with the output stage described above with reference to  FIGS. 5 and 6  have the benefit that, during power delivery into a low impedance, the voltage across the reservoir capacitor  60  decreases progressively after the instant of generator activation. The capacitor value is chosen to be sufficiently large to ensure that the low to high load impedance transition occurring at the start of a tissue cutting or vaporisation cycle can be produced in a single burst of RF energy. Typically, the amount of energy delivered during the initial burst is about 1 joule in a dry environment and between 10 to 20 joules in a wet field environment. The actual energy in the RF pulses or bursts is controlled by the threshold or thresholds set in the voltage sensing circuit  62 , specifically by the difference in supply voltage between pulse initiation and pulse termination. Since the output stage has a very low output impedance, this difference voltage is apparent as a change in delivered RF voltage at the output. The capacitor  60  is, therefore, made sufficiently large (in this embodiment 4.7 mF) that the change in voltage represents only a minor proportion of the absolute voltage at the output. Thus, if the delivered output voltage is a sine wave with a peak voltage of 500V, the supply voltage on supply rail  58 , the size of the capacitor  60  and the transformer TR 1  step-up ratio are chosen such that the output voltage drops by no more than 100V peak (20 percent) during an RF burst. In this preferred embodiment, the output voltage drop is about 50V peak or 10 percent. 
   One of the effects of preventing the supply of lower voltages to the output is that, in a tissue cutting or vaporisation tissue cycle, the voltage is not allowed to drop to a level at which undesirable coagulation effects occur. 
   The preferred generator in accordance with the present invention allows the DC energy fed to the reservoir capacitor  60  to be altered so that the time period during which a cutting voltage is present at the output can be altered. In practice, owing to the low output impedance of the generator, this time period is directly proportional to the stored energy. 
   The control methodology, whereby RF energy bursts or pulses are controlled according to voltage thresholds sensed across a reservoir capacitor, allows very low duty cycles to be used, permitting tissue cutting or vaporisation at low average powers. Indeed, it is possible to operate with less than 5 watts average power (averaged over several capacitor charging and discharging cycles). Accordingly, the generator has uses in low power as well as high power applications. 
   An alternative generator for use in the system described above with reference to  FIG. 1  will now be described with reference to  FIG. 7 . This generator has a variable frequency RF source including a voltage controlled oscillator (VCO)  40 A. In this example, the VCO feeds a divide-by-two stage  40 B which, in turn, feeds a power driver stage  44 A driving an RF output stage in the form of a power bridge  44 B. The power bridge  44 B feeds a resonant output network  80  which delivers a generator output signal across output terminals  74 . In practice, the power driver stage  44 A and the power bridge  44 B can have the same configuration as the power mosfet and driver circuit  44  of the generator described above with reference to  FIG. 3 . The power bridge  44 B takes its DC supply from the supply rail  58  of the DC power supply  50 , but the driver stage  44 A has a lower voltage supply. Typical supply voltages are 180V maximum for the power bridge  44 B and 16.5V for the driver stage  44 A. 
   To bring the frequency of the combination of the VCO  40 A and divide-by-two stage  40 B to the resonant frequency of the output network  80 , the above-described components of the RF source are coupled in a phase-locked loop including a phase sensing element  82  coupled between the power bridge  44 B and the output network  80  to sense the current phase in the input leads to the output network. This current phase signal is applied to one input of a phase comparator  84 , the other input of which receives a signal representative of the output of the VCO  40 A, derived from the output of the divide-by-two stage  40 B via a delay stage  86  which compensates for the delay to the RF signal as it passes through the power driver and the power bridge. 
   As in the first-described generator, the RF output stage  44 B is supplied from the DC supply rail  58  attached to the reservoir capacitor  60 , which allows large currents to be drawn by the output stage  44 B for short periods of time, i.e. currents significantly larger than the current rating of a power supply (not shown) connected to the DC supply rail  58 . It follows that the voltage on supply rail  58  will fall during the time that a large current is drawn. Such variations in voltage are sensed by the voltage sensing stage  62  coupled to the rail  58 . Voltage sensing circuit  62  has a control output coupled to the first transmission gate  64  in a line  88  coupling the divided-down output of the VCO  40 A to the input of the power driver  44 A. 
   As before, the arrangement of the voltage sensing stage  62  and the gate  64  are such that when the voltage on supply rail  58  (the voltage supplied to the power bridge  44 B) drops below a predetermined voltage threshold, the gate  64  is operated to interrupt the signal path between the VCO and the power driver  44 A. When the supply rail voltage rises again, the gate  64  reverts to its conducting state. This may happen when the voltage rises above the threshold mentioned above, or a second threshold voltage. 
   The second transmission gate  72 , connected in series in the signal line  88  with the voltage-operated gate  64 , has a control input connected to the output of a 0.5 second monostable  70  which is triggered by current sensing circuitry comprising the current sensing element  46  in one of the input leads to the output network  80  and the comparator  66 . These elements act to interrupt the signal line  88  to the power driver  44 A for 0.5 seconds when the power bridge output current exceeds a predetermined threshold. 
   Referring to  FIG. 8 , the resonant output network  80  comprises the series combination of an in-line inductance L and a tank capacitor C 1 . The output is taken from across the tank capacitor C 1  (which takes out switching noise) via a first coupling capacitor C 2 . This first coupling capacitor C 2  couples to the output (represented by terminals  74 ) via a step-up matching transformer T with a 1:2 step-up ratio. The secondary rewinding of the transformer T couples to the output terminals via a second coupling capacitor C 3 . In this embodiment, L is about 0.47 μH, the tank capacitor is about 10 nF and the two coupling capacitors C 2  and C 3  co-operate (one of them via the transformer T) to form a coupling capacitance of about 23 nF. 
   It will be appreciated that when the output terminals  74  are open-circuit, the resonant frequency of the output network is determined by the series combination of L and C 1 . When the output terminals  74  are shorted, the resonant frequency is determined by the series combination of L and the network represented by C 1 , C 2 , C 3  and T. With the values given, the short-circuit resonant frequency is about 0.55 times the open-circuit resonant frequency. 
   One of the features of a series-tuned output stage is that peak power delivery inherently occurs at extremely low and extremely high impedances. Referring to  FIG. 9 , the load curve of a series-tuned network (i.e. the delivered power versus load impedance) at resonance is shown by the dotted curve A. The network  80  has minimum power delivery, which may be regarded as the “matched condition”, at a load impedance across the terminals  74  ( FIGS. 7 and 8 ) of about 200 ohms. It will be noted that the part of the curve A which has a negative slope follows a path which is approximately hyperbolic over a major part of its length, which means that this part of the curve is of similar shape to a constant voltage line on the graph of  FIG. 9 . 
   The applicant has recognised that such a characteristic, when applied to the output stage of an electrosurgical generator, allows output power to be maximised for a given constant voltage limit over a range of load impedances. It has been found that erosion of the active electrode of an electrosurgical instrument operated in a conductive liquid increases markedly when the output voltage rises above a threshold in the region of 900 volts to 1100 volts peak-to-peak. By arranging for the load curve of the output network  48  to follow an approximate constant voltage curve at about 1000 volts peak-to-peak (340 volts rms) the power delivered into a varying load impedance can be close to the maximum theoretically achievable for that voltage. 
   In effect, over the range of load impedances of importance in so-called “underwater” electrosurgery, the generator can be made to behave as a constant voltage supply. This can be achieved with a matched output impedance much higher than the load impedance presented by the electrode assembly shown in  FIGS. 2A and 2B  in the wetted condition, which, for a 4 mm loop is in the region of 25 ohms. This translates to a maximum power of about 4.5 kW at 340 volts rms. 
   The actual load curve achieved with the arrangement shown in  FIGS. 7 and 8  is shown by curve B in  FIG. 9 . This deviates from the series-tuned curve A at low impedances owing to imposition of a current limit using the current sensing stage circuitry  46 ,  66  monostable  70  and transmission gate  72  ( FIG. 7 ). In the present embodiment, the current limit is set at a level of about 13 amps. The actual load curve B also deviates from the inherent series-tuned load curve A towards the lower part of the negative-slope portion of the curve A so that the delivered power follows a continuing negative gradient as the load impedance rises, again mimicking a constant voltage supply. This latter deviation is deliberate inasmuch as extreme power into a very high impedance is undesirable. The reason for this deviation is the movement of the resonant frequency of the output network  80 , as described above, coupled with the imposition of a high-frequency limit on the RF frequency output as will be described below. The phase comparator  84  compares the current phase at the input to the output network  80 , as sensed by the phase sensing circuit  82  with a delayed version of the output of the divide-by-two circuit  40 B which, in turn, is fed by the VCO  40 A. Accordingly, the phase and frequency of the VCO are varied to maintain a constant phase at the input to the output network  80 , subject to the upper frequency limit mentioned above. In the absence of other influences, therefore, the output network  80  is maintained in resonance as the load impedance varies. 
   Given that the free-running frequency of the phase-locked loop is arranged to be its maximum frequency of operation, the locking characteristics of the phase-locked loop are such that it can be brought into a locked condition at the minimum frequency, corresponding to minimum load impedance, sufficiently quickly to achieve resonance in the early part of the output pulse, but not so quickly that the current limit circuit (sensing circuitry  46 ,  66  monostable  70  and gate  72 ) fails to trip when the current exceeds a predetermined current threshold. 
   If, now, the output carrier frequency is limited to a value below the frequency of the matched load resonant condition, the delivered power will fall off as the load impedance increases and the resonant frequency correspondingly rises. In fact, the free-run output frequency of the phase locked loop containing the VCO  40 A ( FIG. 7 ) is designed to be this maximum frequency. This ensures that the output network always represents a higher source impedance than the impedance of the load, which affords over-voltage protection in the event of a short. 
   Summarising, to achieve optimum resonant frequency, the excitation oscillator (VCO) is phase-locked to the resonant output network. Defining the range of the VCO provides load curve definition in that the delivered output power falls below the theoretical maximum when the output network resonant frequency rises above the maximum frequency of the divided down output of the VCO  40 A. In other words, a match at high load impedance is prevented by preventing the VCO from generating the higher frequencies necessary for resonance. It also follows that, at high load impedances, the maximum output voltage is controlled by virtue of frequency. 
   It will be seen from  FIG. 9  that the delivered output power is in excess of 1 kW over a range of load impedances corresponding to a wetted or partly wetted electrode. Once vaporisation and arcing has been initiated, the impedance rises, and the delivered power falls. To maintain the average output power at 200 W or less, the output signal is pulsed when the load impedance is low. It will be understood that with a peak power in excess of 4 kW, the pulse duty cycle needs to drop to a level in the region of 5% or less. The pulse repetition rate should be between 5 Hz and 2 kHz, and is preferably at least 10 Hz. These figures are chosen in view of the time taken to initiate vaporisation at the electrode surface. This means that the pulses have a maximum length of about 4 or 5 ms into a low impedance requiring maximum power. Typically, the pulse length is in the region of 1 to 2 ms. While it is not essential, configuring the RF output stage of the generator as an amplifier amplifying the output of a signal derived from a separate oscillator, rather than having a self-oscillating output stage, is preferred in order that full peak power can be achieved within the above-stated pulse lengths. (In this embodiment, the output stage  44 B is an amplifier configured as a power switching bridge for high efficiency.) Should the VCO fail to operate at a frequency corresponding to resonance of the output network  80 , as may happen at the start of each pulse, excessive output currents associated with such a mismatch are prevented since the series-tuned output network is low impedance only at resonance. 
   Pulsing of the output signal can be performed in a number of ways, including simply pulse modulating with predetermined pulse lengths and pulse repetition rates. In the mode of operation of the alternative generator described here, the output is pulsed only during an initial period from the commencement of treatment, the output signal being a continuous wave (CW) signal thereafter, i.e. generally when vaporisation and arcing have been achieved and the load impedance is in an upper range. The duration of the initial period may be fixed or it may be determined by monitoring the load impedance and terminating the initial period when the impedance exceeds a predetermined value. In this embodiment, the duration of the initial period and the length and frequency of the pulses are dynamically variable in response to delivered energy, as measured by the supply rail voltage on supply rail  58 . As has been explained above, high instantaneous power levels are achieved only by allowing the output stage  44 B to draw current from a charge reservoir, here a large capacitance such as capacitor the 47 mF capacitor  60 . As charge is drawn from the capacitor  60 , the supply rail voltage drops. Between pulses, the supply rail voltage rises again. Accordingly, by using gate  64  alternately to allow and prevent the passage of an RF signal along signal line  88  to the power driver  44 A according to the relationship between the supply voltage level and a threshold or thresholds set in the voltage sensing circuit  62 , the output of the generator can be pulsed to achieve maximum peak delivered power whilst operating within a predetermined average power limit. This equilibrium of power consumption and DC supply voltage is achieved by setting the voltage thresholds so that the RF output stage is activated when the supply rail voltage is sufficient to achieve a maximum vaporisation voltage (e.g. 340V rms) and switched off when a lower threshold is reached. The lower threshold defines the maximum energy per pulse and the repetition rate for a given average power level. The initial period referred to above is terminated when the electrode has “fired-up”, in other words when vaporisation and arcing have commenced, so that the load impedance rises and the supply rail voltage stays above the switching threshold or thresholds. In this way it is possible to achieve vaporisation of the conductive liquid surrounding the electrode at impedances as low as 20 ohms without unacceptable erosion of the electrode surface.

Technology Classification (CPC): 0