Patent Abstract:
A method of controlling noise in a pulse width modulation circuit includes varying a sample frequency and a range of information levels, wherein each sample within a data sample stream at the sample frequency represents a level within the range of information levels, to shift in frequency noise generated at the sample frequency during encoding of the data sample stream into pulse width modulated patterns.

Full Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
     This application for patent is a divisional of pending application Ser. No. 11/233,961, filed Sep. 23, 2005. 
     This application is a Continuation-In-Part application of pending application U.S. Ser. No. 10/715,950 entitled “Circuits and Methods for Reducing Interference from Switched Mode Circuits” filed on Nov. 18, 2003, which is a continuation of U.S. Ser. No. 09/651,821 entitled “Circuits and Methods for Reducing Interference from Switched Mode Circuits” filed on Aug. 30, 2000 which is abandoned. 
    
    
     FIELD OF INVENTION 
     The present invention relates in general to switched—mode circuit techniques, and in particular, to circuits and methods for controlling switching noise in switched—mode circuits. 
     BACKGROUND OF INVENTION 
     Class D audio power amplifiers (APAs) have been used for many years in systems, such as wireline telephony, where high bandwidth is not critical. More recently however, new fabrication techniques, and in particular, new techniques for fabricating power transistors, have made integrated Class D APAs possible. This result has extended their potential applications to lower-power, higher-bandwidth systems, including battery-powered portable music players and wireless communications devices. 
     One major advantage of Class D amplifiers is their efficiency. Generally, an input signal is converted into a relatively high frequency stream of pulses varying in width with the amplitude of the input signal. This pulse width modulated (PWM) signal switches a set of power output transistors driving an output load between cutoff and saturation, which results in efficiencies above ninety percent (90%). In contrast, the typical Class AB push-pull amplifier, using output transistors in which their conduction varies linearly during each half-cycle, has an efficiency of only around sixty percent (60%). The increased efficiency of Class D amplifiers in turn reduces power consumption and consequently lowers heat dissipation and improves battery life. 
     Similarly, switched mode power supplies have found wide acceptance in the design of compact electronic appliances. Switched mode power supplies advantageously use smaller transformers and are therefore typically more compact and of lighter weight. These features are in addition to the increased efficiency realized over linear power supplies. Moreover, the total number of components can be reduced to, for example, a power MOSFET die and a PWM controller die packaged together in a single package. 
     One of the disadvantages of using conventional switched mode devices is the interference (radiated and conducted) generated by the switching mechanism. This problem is of particular concern in compact electronic appliances which include a radio and similar audio circuits. For example, if the switching frequency of the given switched—mode device is nominally at 350 kHz, harmonics will be generated at 700 kHz, 1050 kHz and 1400 kHz, all of which fall within the AM radio broadcast band. 
     Given the importance of improved battery-life, reduced heat dissipation, and component size minimization in the design and construction of portable electronic appliances, improved switched mode techniques will have numerous practical advantages. The possible applications for these techniques are numerous, although Class D APAs and switched mode power supplies are two primary areas which should be considered. 
     SUMMARY OF INVENTION 
     The principles of the present invention are embodied in circuits and methods for controlling interference in systems, such as radio receivers, utilizing switched-mode circuitry. According to one particular embodiment of these principles, a method is disclosed for controlling noise in a pulse width modulation circuit, which includes varying a sample frequency and a range of information levels, wherein each sample within a data sample stream at the sample frequency represents a level within the range of information levels, to shift in frequency noise generated at the sample frequency during encoding of the data sample stream into pulse width modulated patterns. 
     The inventive concepts address one of the major disadvantages of conventional switched mode devices, namely, interference (noise) caused by the switching mechanism itself. This interference is of particular concern in systems employing radio receivers and similar interference sensitive circuitry. In accordance with the inventive principles, the switching frequency is shifted as a function of the radio frequency being received such that the switching frequency and its harmonics fall outside the frequency band of the received signal. Advantageously, these principles can be applied to different types of switched circuitry, including Class D amplifiers and power supplies. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
       For a more complete understanding of the present invention, and the advantages thereof, reference is now made to the following descriptions taken in conjunction with the accompanying drawings, in which: 
         FIG. 1  is a high-level block diagram of a representative radio receiver suitable for describing one particular application of the principles of the present invention; 
         FIG. 2A  is a block diagram of an exemplary audio power amplifier (APA) embodying the principles of the present invention and suitable for utilization in the radio receiver of  FIG. 1 ; and 
         FIG. 2B  is a more detailed block diagram of the pulse-width modulation (PWM) stage shown in  FIG. 2A . 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     The principles of the present invention and their advantages are best understood by referring to the illustrated embodiment depicted in  FIGS. 1-2  of the drawings, in which like numbers designate like parts. 
       FIG. 1  is block diagram of one channel of a digital radio  100  embodying the principles of the present invention. Digital radio  100  includes an analog section or front-end  101  which receives radio frequency (RF) signals from an associated antenna  102 . Analog front-end  101  is preferably a conventional RF down-converter including a low noise amplifier (LNA)  103  for setting the system noise figure, a bandpass filter  104 , and a mixer  105  driven by an analog local oscillator  106 . The mixed-down analog signal is then converted into digital form by an analog to digital converter  107 . 
     The digitized data output from A/D converter  107  is passed to digital processing section  108 . Mixers  109   a  and  109   b  generate in-phase (I) and quadrature (Q) signals from a corresponding pair of clock phases from crystal oscillator  110 . The I and Q signals are next passed through bandpass filters  111   a  and  111   b  and on to digital baseband processor  112 . The processed digital signal is then re-converted to analog (audio) form by D/A converter  113 . 
     According to the principles of the present invention, a switched-mode (Class D) audio power amplifier (APA)  114 , discussed in detail below, is used to drive an external set of speakers or a headset (not shown). Preferably, at least some of the components of digital radio  100  are powered by a switched-mode power supply (SMPS)  114 . 
     As discussed above, Class-D amplifiers generally utilize pulse-width (duty cycle) modulation (PWM) techniques, which realize a high efficiency amplifier operation by directly pumping electric current into an inductor-capacitor (L-C) filter through low-resistance power transistors. These low-resistance transistors are typically driven by a delta-sigma modulator, which quantizes and noise shapes the input signal, and a pulse width modulation (PWM) stage, which encodes the quantized samples output from the delta-sigma modulator into a stream of PWM encoded patterns. The output from the PWM stage drives the gates of the power transistors, which may be in either a half-bridge or full bridge configuration. The fundamental switching frequency of the power transistors approximates the output sample frequency of the delta-sigma modulator quantizer, F SO . 
     A convenient system design, often utilized in audio systems, is based on the relationship:
 
 F   MCLK   =F   SO ·( N   Level −1)
 
in which F MCLK  is the master clock signal (oversampling) frequency of the system, N Level  is the number of the quantizer levels at the delta-sigma modulator quantizer output, and F SO ·(N Level −1) is the data or “chip” rate of the PWM encoded patterns output from the PWM encoder.
 
     In digital radio systems including a Class D amplifier operating at a typical output sample frequency F SO  of several hundreds of KHz, the resulting switching related interference can make it very difficult to tune to radio stations operating at a frequency at, or close to, the F SO  frequency. One possible solution to addressing this problem is to shift the master clock frequency F MCLK , and hence the output sample frequency F SO , when tuning to a radio station broadcasting at the same or a close frequency. However, this technique is costly to implement as it generally requires utilization of a phase-locked loop (PLL) circuit in the system. 
     According to the principles of the present invention, the master clock signal frequency F MCLK  remains fixed, while the sample output frequency F SO  is varied during radio frequency tuning to avoid the radio frequency band of interest. At the same time, the number of quantizer levels N Level  is varied to preserve the information content in the PWM encoded signal. Generally, when the output sample frequency F SO  is reduced, the number of quantizer levels N Level  increases, and vice versa. 
       FIG. 2A  is more detailed block diagram of APA  114  shown in  FIG. 1 . Generally, the embodiment of APA  114  shown in  FIG. 2A  is based on a pulse width modulation (PWM) stage  200 , which is shown in further detail in  FIG. 2B . PWM stage  200  includes a sample rate converter (SRC)  201 , which converts a stream of digital samples from the input sample frequency F SI  to the digital output sample frequency F SO . The digital data stream output from SRC  201  is input into a delta-sigma modulator  202 , which performs noise shaping and requantizes the data stream into requantized digital data samples, each representing N-number of information levels, at the output sample frequency F SO . The requantized samples generated by delta-sigma modulator  202  are provided to a PWM encoder  203 , which generates PWM encoded data patterns each representing one of the N-number of levels defined by the corresponding requantized data sample. PWM signal generation techniques are discussed in co-assigned U.S. Pat. No. 5,812,102 to Melanson, entitled “Delta Sigma PWM DAC to Reduce Switching,” and incorporated herein by reference. The PWM encoded data drive a set of power output transistors  220 , which may be in either a full-bridge or half-bridge configuration. 
     The number of levels N Level  represented by each sample output from delta-sigma modulator  202  into PMW encoder  203  generally depends on the frequency at which radio  100  is tuned to, the input data frequency F SI , and the output frequency F SO  of the data stream output from SRC  201 . Generally, the number of levels N Level  varies as the output sample frequency F SO  “hops” to avoid the frequency at which radio  100  is being tuned. For discussion purposes, a two—frequency hopping system based on the following characteristics will be assumed, although the number of possible frequency hops in the output sample frequency F SO , as well as the chosen master clock frequency F MCLK , the chosen output sample frequencies F SO , and the number of quantizer levels, may vary depending on the specific system. The base mode for this exemplary embodiment is:
 
F MCLK =24.576 MHz
 
F SO =384 kHz
 
N Level =65.
 
For the frequency-shift (frequency hopping) mode of operation:
 
F MCLK =24.576 MHz
 
F SO =341.333 kHz
 
N Level =73.
 
     It is a common practice to design a quantizer, such as the quantizer of delta-sigma modulator  202 , to output quantized samples representing 2 n +1 number of information levels because a 2 n +1 level quantizer typically performs a simple truncation, which eliminates all but the first n+1 number of most significant bits (MSBs) of the sample. On the other hand, changing the quantization level to 73, for example, requires a 73-level quantizer, which cannot be implemented utilizing simple truncation. However, one economical way of implementing 73-level quantization, is to multiply each sample by 72 and then extract the first 7 MSB&#39;s by truncation. Advantageously, 72 can be numerically decomposed as 9×8 so that, in actual computation, a multiplication by 72 only amounts to a multiplication by 9 followed by a shift by 3 MSBs. Furthermore, because 9 is equal to 8+1 and a multiplication by 8 only requires a fixed shifter, a multiplication by 9 can also be realized by a simple adder and a shifter. 
     When a quantizer is used in the context of delta-sigma modulator, there always is a feedback path from the quantizer output to a summer at the modulator input. Consequently, if the data samples are multiplied by 9 before quantization, the quantizer output must be divided by 9 before feedback. In some delta-sigma modulator topologies, mostly “feedforward” architectures, an actual division (or, equivalently, a multiplication) must be performed in the feedback loop during frequency hopping. In other topologies, mostly “feedback” architectures, however, the division can be done implicitly by simply scaling the feedback coefficients during frequency hopping by 9 using a read-only memory (ROM) based table. 
     As shown in  FIG. 2A , the system master clock (MCLK) signal is provided to an MCLK frequency divider  204 . In the illustrated embodiment, PWM stage  200  operates at one of two output sample frequencies F SO , namely F MCLK /64 and F MCLK /72. Consequently, two corresponding clock signals at the frequencies F MCLK /64 and F MCLK /72, along with a clock signal at the frequency F MCLK /64 /8 and a clock signal at the frequency F MCLK /72 /9, both of which corresponds to a frequency of eight (8) times the corresponding output sample frequency F SO , are output by MCLK frequency divider  204 . 
     Sample clock signal recovery circuitry  205  receives the sample clock signal ( LRCK ) at the input sampling frequency F SI  and outputs internal clock signals at selected multiples of F SI . In the illustrated embodiment, sample clock signal recovery circuitry  205  generates internal clock signals of eight times the input sampling frequency F SI  (8*F SI ), sixteen times the input sampling frequency F SI  (16*F SI ), one hundred and twenty eight times the input sampling frequency F SI  (128*F SI ), and two hundred and fifty six times the input sampling frequency F SI  (256*F SI ). Sample clock signal recovery circuitry also determines whether the input data stream will be interpolated by two in ×2 interpolator  206  or input directly at the input sample frequency F SI . 
     Shifting the number of PMW levels, as required to avoid interference in the radio frequency band is implemented by the PWM level select ( PWML ) signal generated in response to user tuning of radio  100  of  FIG. 1 . In response to the  PWML  signal, a selector  207  selects between the clock signal at the frequency F MCLK /64 /8 or the clock signal at the frequency F MCLK /72 /9, as generated by MCLK frequency divider  204 . At the same time, in response to the interpolation decision performed by sample clock signal recovery circuitry  205 , multiplexer  208  selects from either the clock signal at the frequency 8*F SI  or the clock signal 16*F SI  generated by sample clock signal recovery circuitry  205 . The selected signal output by multiplexer  207 , and the clock signal selected by multiplexer  208 , are provided to upsample rate estimator  209 , which generates an output representing the output sample frequency to input sample frequency ratio F SO /F SI . 
     The  PWML  signal also controls the selection by multiplexer  210  of either the clock signal at the frequency F MCLK /64 or the clock signal at the frequency F MCLK /72 generated by MCLK frequency divider  204 . Concurrently, the interpolation decision signal generated by sample clock signal recover circuitry  205  selects between the clock signals at the frequencies 128*F SI  and 256*F SI , generated by sample clock signal recovery circuitry  205 , through a multiplexer  211 . The clock signal at the output frequency F SO  selected by multiplexer  210  and the clock signal selected by multiplexer  211  provided to a rate estimator circuit  212 . 
     In response to the frequency ratio F SO /F SI  generated by upsample rate estimator  209 , the current output frequency F SO  selected by multiplexer  210  in response to the  PWML  signal, and the oversampling rate selected by multiplexer  211 , rate estimator  212  provides an estimate of the difference between the output sample frequency F SO  and the input sample frequency F S . The estimation provided by rate estimator  212  is utilized by SRC polynomial evaluation circuitry  215 , which controls the selection of the coefficients required by SRC  201  of  FIG. 2B  to convert the input sample frequency F SI  of the data stream output from first and first out (FIFO) register  216  to the output sample frequency F SO . The  PWML  signal additionally selects a set of coefficients from read only memory (ROM)  214 , which are utilized by delta-sigma modulator  202  of  FIG. 2B  for noise shaping the sample stream at the selected output sample frequency F SO . 
     In alternate embodiments of radio receiver  100 , the source sample rate may be varied in ADCs  107  to implement frequency hopping. In these embodiments, SRC  201  of  FIG. 2A  may be eliminated. 
     The same inventive principles may be similarly implemented in switched-mode power supply  115  of  FIG. 1  to ensure that any generated switching noise is shifted out of the radio frequency band of interest to avoid interference during signal reception by radio  100 . In sum, according to the inventive principles, the switching frequency of a switched-mode (PWM) circuit is shifted such that the fundamental switching frequency and its harmonics fall outside the frequency band of interest, while the quantization level of the generating digital sample stream is proportionally varied to minimize the loss of information. 
     Although the invention has been described with reference to specific embodiments, these descriptions are not meant to be construed in a limiting sense. Various modifications of the disclosed embodiments, as well as alternative embodiments of the invention, will become apparent to persons skilled in the art upon reference to the description of the invention. It should be appreciated by those skilled in the art that the conception and the specific embodiment disclosed might be readily utilized as a basis for modifying or designing other structures for carrying out the same purposes of the present invention. It should also be realized by those skilled in the art that such equivalent constructions do not depart from the spirit and scope of the invention as set forth in the appended claims. 
     It is therefore contemplated that the claims will cover any such modifications or embodiments that fall within the true scope of the invention.

Technology Classification (CPC): 7