Patent Abstract:
A charge pump circuit includes a high-swing transconductance amplifier. A high input swing transconductance is provided in a negative feedback loop of the charge pump circuit without an abrupt change in transconductance. The high-swing transconductance amplifier includes a transconductance cell and high-swing circuitry. The transconductance cell includes a current supply transistor, which provides current for transconductance while input voltages are within the operational range for the transconductance cell. When the input voltages increase so as to be outside of the operational range, the current source transistor enters into triode region of operation, and provides reduced current. The high-swing circuitry supplies the current in this case so that abrupt change in transconductance does not occur. The high-swing circuitry widens the output compliant voltage range of the charge pump circuit and hence reduces the sensitivity requirement of the VCO, Kvco, in any PLL design, in particular design for PLLs used in tuners.

Full Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
   The present application claims priority of U.S. Provisional Patent Application No. 60/235,725 entitled “High-Swing Transconductance Amplifier for Charge Pump Circuit,” filed Sep. 27, 2000, the contents of which are hereby incorporated by reference in full. 
   The present applications contains subject matter related to the subject matter disclosed in commonly owned U.S. Published Patent Application 2002/0,050,861 A1 (“the &#39;861 PPA”) entitled “Variable Transconductance Variable Gain Amplifier Utilizing a Degenerated Differential Pair,” published May 2, 2002, the contents of which are incorporated by reference in full. 

   FIELD OF THE INVENTION 
   The present invention is related to charge pump circuits, and particularly to a high-swing transconductance amplifier for providing transconductance over an increased range of input voltages. 
   BACKGROUND OF THE INVENTION 
   Phase-locked loop (PLL) circuits are used in various different applications including, but not limited to, frequency tuners (e.g., satellite tuners) for selecting different television and/or radio channels. In a PLL circuit, a feedback loop is generally used to adjust frequency/phase of a voltage-controlled oscillator (VCO) output signal until the VCO output signal aligns with a reference clock signal. 
   The PLL circuit typically includes a charge pump circuit to provide voltage control signal Vc with sufficient charge for proper VCO operation. The charge pump circuit typically includes a transconductance cell, which may also be referred to as a transconductance amplifier, to generate current using input voltage signals, which are adjusted based on phase/frequency relationship between the reference clock and the VCO output signal. 
   The voltage control signal Vc output by the charge pump circuit is often limited by the transconductance capability of the transconductance cell. For example, the transconductance cell typically includes a current supply transistor that enters into triode region of operation as the voltage control signal Vc increases, resulting in abrupt change to the transconductance capabilities since the current supply transistor typically provides less current when operating in triode region. The sensitivity requirement of the VCO, Kvco, in any PLL design may be reduced when the VCO control voltage range of the charge pump circuit can be widened. 
   Therefore, it is desirable to provide a transconductance cell that is capable of transconductance over a wider range of input voltages. 
   SUMMARY OF THE INVENTION 
   In one embodiment according to the present invention, a transconductance amplifier is provided. The transconductance amplifier includes a transconductance cell for receiving one or more input voltage signals and for generating one or more first currents using the input voltage signals. The transconductance cell is used to supply a current output including at least a portion of the first currents. The transconductance amplifier also includes high-swing circuitry for receiving the input voltage signals, for generating one or more second current signals using the input voltage signals and for providing at least a portion of the second current signals to the transconductance cell to be included in the current output. The first currents provide more than half of the current output while the input voltage signals are within first range of voltages. The second currents provide more than half of the current output while the input voltage signals are not within the first range of voltages. 
   In another embodiment according to the present invention, a method of generating a current output using a transconductance amplifier is provided. The transconductance amplifier includes a transconductance cell and high-swing circuitry. One or more input voltage signals are received. One or more first currents are generated in the transconductance cell for inclusion in the current output. One or more second currents are generated in the high-swing circuitry for inclusion in the current output. At least a portion of the second currents is provided to the transconductance cell for inclusion on the current output. The first currents provide more than half of the current output while the input voltage signals are within first range of voltages. The second currents provide more than half of the current output while the input voltage signals are not within the first range of voltages. 
   In yet another embodiment according to the present invention, a charge pump is provided. The charge pump includes an i/o circuit for receiving one or more voltage difference signals and for generating a voltage control signal and one or more input voltage signals. The charge pump also includes a transconductance amplifier having a transconductance cell and high-swing circuitry. The transconductance amplifier is used to provide a current output to the i/o circuit, and the current output is used to provide charge for the voltage control signal. The voltage control signal and the input voltage signals are generated based on the voltage difference signals. The transconductance cell generates more than half of the current output when the input voltage signals are within a first range of voltages. The high-swing circuitry generates more than half of the current output when the input voltage signals are not within the first range of voltages. 
   In still another embodiment of the present invention, a phase-locked loop (PLL) is provided. The PLL includes a phase detector for receiving a reference clock signals and a voltage controlled oscillator (VCO) output signal, and for generating one or more voltage difference signals. The PLL also includes a charge pump for receiving the voltage difference signals and for generating a voltage control signal with sufficient charge for VCO operation. In addition, the PLL includes a VCO for receiving the voltage control signal and for generating the VCO output signal based on the voltage control signal. The charge pump includes a transconductance cell and a high-swing circuitry. The transconductance cell generates more than half of the charge when the VCO output signal is within a first range of voltages. The high-swing circuitry generates more than half of the charge when the VCO output signal is not within the first range of voltages. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     These and other aspects of the invention may be understood by reference to the following detailed description, taken in conjunction with the accompanying drawings, which are briefly described below. 
       FIG. 1  is a block diagram of a phase-locked loop (PLL) circuit, which may include an embodiment according to the present invention; 
       FIG. 2  is a block diagram of a charge pump including a charge pump I/O circuit and a transconductance cell; 
       FIG. 3  is a circuit diagram of the charge pump of  FIG. 2 ; 
       FIG. 4  is a block diagram of a charge pump according to an embodiment of the present invention; 
       FIG. 5  is a partial circuit diagram of a charge pump of  FIG. 4 ; 
       FIG. 6  is a circuit diagram of a high-swing transconductance cell according to an embodiment of the present invention; and 
       FIGS. 7 and 8  illustrate a flow diagram of the process of providing currents from the high-swing circuitry to the transconductance cell of  FIG. 6 , such that the current provided to the charge pump I/O circuit is not abruptly decreased when the tail current source enters into the triode region of operation. 
   

   DETAILED DESCRIPTION 
     FIG. 1  is a block diagram of a phase-locked loop (PLL) circuit  100 , which may include an embodiment according to the present invention. The PLL circuit  100  includes a phase detector  104 , a loop filter  106 , a charge pump  108 , and a voltage controlled oscillator (VCO)  110 . The phase detector  104  preferably receives a reference clock  102  and an output  112  of the VCO  110 . The VCO output  112  is an output of the PLL circuit  100 , and is fed back into the PLL circuit. 
   The phase detector  104  compares phase and/or frequency between the reference clock  102  and the VCO output  112 . The phase detector  104  preferably generates one or more signals to indicate phase and/or frequency difference between the VCO output  112  and the reference clock  102 , and provides the difference signals to the charge pump  108  via the loop filter  106 . The loop filter  106  may be a low pass filter used to filter out high frequency components of the difference signals from the phase detector  104 . 
   The charge pump  108  preferably accumulates electrical charge based on the difference signals and provides a voltage control signal with sufficient current to the VCO  110  to adjust phase and/or frequency of the VCO output  112 . The charge pump  108  preferably includes a transconductance cell for generating current using input voltages based on the difference signals from the phase detector  104 . The charge pump  108  in one embodiment of the present invention may include high-swing circuitry for increasing the range of input voltages over which the transconductance of the transconductance cell does not change abruptly. 
   The PLL circuit of  FIG. 1  is shown for illustrative purposes only. The present invention may be applied to any PLL circuit as well as any other circuit including a charge pump. For example, the charge pump of the present invention may be applied to the PLL circuit of FIG. 17, the frequency synthesizer of FIG. 18, and a PLL 4506 of FIG. 46 a , all of which are disclosed in the above mentioned &#39;861 PPA the contents of which have been incorporated by reference in full. 
     FIG. 2  is a block diagram of a charge pump  150 . The charge pump  150  includes a charge pump input/output (I/O) circuit  152  and a transconductance cell  154 . The transconductance cell  154  may also be referred to as a transconductance amplifier. The charge pump  150 , for example, may be used in the charge pump  108  of FIG.  1 . 
   The charge pump I/O circuit  152  preferably receives difference signals  156 ,  158 ,  160  and  162  from a phase detector, such as, for example, the phase detector  104  of FIG.  1 . The difference signals include two pairs of differential signals  156 ,  158  and  160 ,  162 . The first pair of differential signals  156 ,  158  includes an UP signal and an {overscore (UP)} signal. The UP and {overscore (UP)} differential signals preferably are generated by the phase detector when the output of the VCO has lower frequency and/or phase lag compared to a reference clock. The UP and {overscore (UP)} differential signals preferably increase frequency of the VCO output and/or phase shift (towards phase lead) the VCO output by increasing a VCO control signal Vc  170  provided to a VCO, such as the VCO  110  of FIG.  1 . The VCO control signal Vc  170  may also be referred to as a control voltage. 
   The second pair of differential signals  160 ,  162  includes a DOWN signal and a {overscore (DOWN)} signal. The DOWN and {overscore (DOWN)} differential signals  160 ,  162  are generated by the phase detector when the output of the VCO has higher frequency and/or phase lead compared to the reference clock. The DOWN differential signals preferably decrease frequency of the VCO output and/or phase shift (towards phase lag) the VCO output by decreasing the VCO control signal Vc  170  provided to the VCO. 
   The charge pump I/O circuit  152  preferably provides voltage difference signals Vap  164  and Vcap  166  to the transconductance cell  154 . The transconductance cell  154  preferably generates current based on the voltage level of the Vap and Vcap signals  164 ,  166 , and provides a negative feedback current  168  to the charge pump I/O circuit  152  to provide sufficient charge to the VCO for its operation. When the PLL is in lock, the VCO control signal Vc  170  and the Vap and Vcap signals  164 ,  166  may all be at the identical level because of the negative feedback loop used. 
     FIG. 3  is a circuit diagram of a charge pump  200 . The charge pump  200 , for example, may be similar to the charge pump  150  of FIG.  2 . The charge pump  200  includes a charge pump I/O circuit  202  and a transconductance cell  204 . 
   The charge pump I/O circuit  202  includes two positive channel (p-channel) metal-oxide semiconductor (PMOS) transistors  206  and  208 . A source of the PMOS transistor  206  is coupled to a voltage supply. Depending on the fabrication technology used, the voltage supply may be 3.3 V, 1.8 V, 1.3 V or any other suitable voltage. Gates of the PMOS transistors  206  and  208  are coupled to a suitable bias voltage supplied by respective biasing circuits (not shown). Design and use of biasing circuits for biasing PMOS transistors  206 ,  208  as well as for biasing various different PMOS and NMOS transistors used for implementation of the present invention are well known to those skilled in the art. 
   A drain of the PMOS transistor  206  is coupled to a source of the PMOS transistor  208 , and a drain of the PMOS transistor  208  is coupled to UP and {overscore (UP)} switches  214  and  210 . Thus, the PMOS transistors  206  and  208  are coupled in series between the voltage supply and the UP and {overscore (UP)} switches. The PMOS transistors  206  and  208  function as a current source for providing p-channel current I p . 
   The UP and {overscore (UP)} switches  214 ,  210  and DOWN and {overscore (DOWN)} switches  216 ,  212  are depicted as switches in  FIG. 3 , but in practice, they may be implemented using transistors. The UP and {overscore (UP)} differential signals and DOWN and {overscore (DOWN)} differential signals preferably are digital signals that take on the value of high or low (e.g., “1” or “0”). For example, when the VCO output lags in phase of and/or is lower in frequency than the reference clock signal, the UP signal of high (or on) is applied to the UP switch  214  while the DOWN signal of low (or off) is applied at the switch  216 . In other embodiments where inverse logic has been used, the UP signal of low and the DOWN signal of high may be applied at the UP switch  214  and the DOWN switch  216 , respectively. 
   The {overscore (UP)} and {overscore (DOWN)} signals are opposite of the UP and DOWN signals, respectively. In this case, when the UP and {overscore (DOWN)} signals are high and the {overscore (UP)} and DOWN signals are low, the switch  214  and the switch  212  are closed while the switches  210  and  216  remain open. When the VCO output leads in phase of and/or is higher in frequency than the reference clock, on the other hand, the {overscore (UP)} switch  210  and the DOWN switch  216  preferably are closed and the UP switch  212  and the {overscore (DOWN)} switch  214  preferably remain open. 
   When the UP switch  214  is closed, the p-channel current I p  preferably flows into a capacitor  220 , and into a capacitor  224  via a resistor  222 . As the capacitors  220  and  224  are charged up, a VCO control signal Vc  234  and the Vcap voltage  232  increase. Meanwhile, the {overscore (DOWN)} switch  212  is closed, and a capacitor  218  is discharged through negative-channel (n-channel) metal-oxide semiconductor (NMOS) transistors  226  and  228 , thus decreasing Vap voltage  230 . A drain of the NMOS transistor  226  is coupled to the DOWN and {overscore (DOWN)} switches  216  and  212 . A drain of the NMOS transistor  228  is coupled to a source of the NMOS transistor  226  and receives a negative feedback current  236 . A source of the NMOS transistor  228  is coupled to ground. Gates of the NMOS transistors  226  and  228  are coupled to suitable bias voltages from respective biasing circuits (not shown), the design and use of which are well known to those skilled in the art. 
   When the DOWN switch  216  is closed, the capacitors  220  and  224  are discharged via the NMOS transistors  226  and  228 . As these capacitors are discharged, the VCO control signal Vc  234  and the Vcap voltage  232  tend to decrease. When the VCO Control signal Vc  234  decreases, the VCO tends to decrease in frequency and/or shifts phase (towards phase lag) to be in line with the reference clock signal. Meanwhile, the {overscore (UP)} switch  210  preferably closes, and the p-channel current I p  preferably flows into the capacitor  218 , tending to increase the Vap voltage  230 . 
   When the PLL is in lock, the VCO control voltage Vc  234  preferably should be kept constant. To keep the VCO control voltage Vc  234  constant, the charge pump  200  should be kept quiet, i.e., the UP and DOWN signals should be kept low. Frequency/phase detectors, such as, for example, the phase detector  104  of  FIG. 1 , are typically designed so that, when the PLL achieves lock, both the UP and DOWN signals become high for a very small fraction of a clock period. Then, both the UP and DOWN signals become low, making the {overscore (UP)} and {overscore (DOWN)} signals high. When both the {overscore (UP)} and {overscore (DOWN)} signals are high, the {overscore (UP)} and {overscore (DOWN)} switches  210  and  212  are closed, the p-channel current I p  flows through them, and a feedback loop is established through the Vap and Vcap voltages. 
   The Vap voltage  230  and the Vcap voltage  232  are provided to the transconductance cell  204  as Vap voltage  254  and Vcap voltage  256 , respectively. The transconductance cell  204  includes a PMOS transistor  238  as a tail current source. The term “tail current source” often designates a current source that is connected from supply (power or ground) to the source of a differential pair of transistors. 
   A source of the PMOS transistor  238  is coupled to the voltage supply, which, for example, may be 3.3V or any other suitable voltage. A gate of the PMOS transistor  238  is coupled to a suitable bias voltage from a biasing circuit (not shown), the design and use of which is well known to those skilled in the art. A drain of the PMOS transistor  238  is coupled to sources of PMOS transistors  240  and  242 . Gates of the PMOS transistors  240  and  242  are coupled to the Vap voltage  254  and the Vcap  256  voltage, respectively. The Vap voltage  254  and the Vcap voltage  256  preferably are identical to the Vap voltage  230  and the Vcap voltage  256 . 
   The Vap voltage  254  and the Vcap voltage  256  preferably are used to control the magnitude of the current supplied by the PMOS transistor  238  (tail current source). The current I 1  through the PMOS transistor  240  flow through a PMOS transistor  244  and then is divided between the negative feedback current  236  and the current through an NMOS transistor  250 . The current I 2  through the PMOS transistor  242  flows through a PMOS transistor  246 , an NMOS transistor  248 , and then through an NMOS transistor  252 . 
   Gates of the NMOS transistors  250  and  252  are coupled to one another and to a drain of the NMOS transistor  248 . A source of the NMOS transistor  248  is coupled to a drain of the NMOS transistor  252 . Sources of the NMOS transistors  250  and  252  are coupled to ground. 
   A portion of the current I 1  is provided to the charge pump I/O circuit as the negative feedback current  236 . The negative feedback current  236  preferably adjusts the n-channel current I n  to match the p-channel current I p . The transconductance cell  204  preferably performs such adjustment of the negative feedback current  236  through detecting the difference in voltages between the Vap voltage  230  and the Vcap voltage  232 . 
   As the Vap and Vcap voltages  254  and  256  increase, the PMOS transistors  240  and  242  tend to shut off, and the currents I 1  and I 2  tend to decrease. The voltage at the source of the PMOS transistors  240  and  242  tend to increase, and Vds voltage between the source and the drain of the PMOS transistor  238  tends to decrease. In this case, the PMOS transistor  238  may enters into triode region of operation. This typically occurs when the Vds of the PMOS transistor  238  is less than the effective voltage at its gate. As the PMOS transistor  238  enters into the triode region of operation, the transconductance drops since the current through the PMOS transistor  238  decreases. 
   In the charge pump  200  of  FIG. 3 , therefore, the upper limit of the control voltage Vc is limited by the input swing of the differential pair of PMOS transistors  240  and  242  in the transconductance cell. In this case, the input swing of the PMOS transistors  240  and  242  is limited by the Vgs between the gate and the source of these PMOS transistors plus the drain saturation voltage (Vdsat) of the PMOS transistor  238  (tail current source). For example, when Vds is bigger than Vdsat, the transistor is generally said to be operating in saturation region, and when Vds is smaller than Vdsat, the transistor is generally said to be operating in triode region. 
     FIG. 4  is a block diagram of a charge pump  151  in an embodiment according to the present invention. The charge pump  151  includes a charge pump I/O circuit  152  and a transconductance cell  154 , which are similar to the corresponding components of the charge pump  150  of FIG.  2 . In addition to the charge pump I/O circuit  152  and the transconductance cell  154 , the charge pump  151  includes high-swing circuitry  155 . The high-swing circuitry  155  and the transconductance  154  form a high-swing transconductance cell  153 , which may also be referred to as a high-swing transconductance amplifier. 
   The high-swing circuitry  155  receives Vap and Vcap voltages  172  and  174  from the charge pump I/O circuit  152 . The high-swing circuitry  155  preferably provides current  176  to the transconductance cell  154  when the tail current source (e.g., PMOS transistor) in the transconductance cell  154  enters into triode region of operation and the output current  168  tends to decrease. This way, the output current  168  provided to the charge pump I/O circuit  152  preferably is maintained even when the tail current source enters into the triode region of operation. Thus, the operational range of the VCO control signal Vc  170  of the charge pump  151  is increased compared to the charge pump  150  (without a high-swing circuitry) of  FIG. 2 , provided that both the charge pump  151  and the charge pump  151  receive the same power supply voltage, e.g., Vdd, from the voltage supply. 
     FIG. 5  is a partial circuit diagram of a charge pump  201  in an embodiment according to the present invention. The charge pump  201  is similar to the charge pump  200  of  FIG. 3  except that the charge pump  201  includes high-swing circuitry  205 . The high-swing circuitry  205  preferably receives Vap and Vcap voltages  262  and  264  from the charge pump I/O circuit  202 . When the transistor  238  enters into triode region of operation, and thus I 1  and I 2  currents tend to decrease, the high-swing circuitry  205  preferably supplies currents  258  and  260  so as to maintain the magnitude of the I 1  and I 2  currents. This way, the negative feedback current  236  supplied to the charge pump I/O circuit  202  does not undergo abrupt changes when the PMOS transistor  238  enters into the triode region of operation. 
     FIG. 6  is a circuit diagram of a high-swing transconductance cell  203  of FIG.  5 . The high-swing transconductance cell  203 , in addition to the transconductance cell  204 , includes high-swing circuitry  205 . The high-swing circuitry  205  preferably maintains I 1  and I 2  currents when the PMOS transistor  238  (tail current source) enters into the triode region of operation so that the negative feedback current  236  provided by the high-swing transconductance cell  203  to the charge pump I/O circuit  202  does not abruptly decrease. 
   The high-swing circuitry  205  includes PMOS transistors  286 ,  288 ,  290  and  292 . A gate and a drain of the PMOS transistor  288  are coupled to one another, and a source of the PMOS transistor  288  is coupled to voltage supply, which may be 3.3V. The voltage supply in other embodiments may be 1.8V, 1.3V or any other suitable voltage for supplying power to the high-swing transconductance cell  203 . Those skilled in the art would appreciate that the selection of the voltage level for the voltage supply typically depends on the type of fabrication technology used to fabricate the circuitry. 
   Since the gate and the drain of the PMOS transistor  288  are coupled to one another and the source is coupled to the voltage supply, the PMOS transistor  288  operates similarly to a diode current source. A source of the PMOS transistor  286  is coupled to the voltage supply and a gate of the PMOS transistor  286  is coupled to the gate of the PMOS transistor  288 . Therefore, the PMOS transistor  286  is configured as a current mirror of the PMOS transistor  288 . In other words, the currents flowing through the PMOS transistors  286  and  288  would be similar in magnitude to each other as long as the PMOS transistors  286  and  288  have similar dimensions. In an embodiment according to the present invention, the PMOS transistors  286  and  288  preferably have similar dimensions. 
   A source of the PMOS transistor  290  is coupled to the voltage supply, and a gate and a drain of the PMOS transistor  290  are coupled to one another. Therefore, similarly to the PMOS transistor  288 , the PMOS transistor  290  is configured as a current source diode. The PMOS transistor  292  has its source coupled to the voltage supply and its gate coupled to the gate of the PMOS transistor  290 . Therefore, the PMOS transistor  292  is configured as a current mirror of the PMOS transistor  290 . 
   Drains of the PMOS transistors  286  and  292  are coupled to the drains of the PMOS transistors  240  and  242 , respectively, over current supply lines  260  and  258 . When the Vap and Vcap voltages  254  and  256  increase so as to force the PMOS transistor  238  into triode region of operation, the PMOS transistors  286  and  292  preferably provide currents over the current supply lines  260  and  258  to make up for the reduction in currents through the PMOS transistors  240  and  242 , respectively. 
   The drains of the PMOS transistors  288  and  290  are coupled to drains of NMOS transistors  294  and  296 , respectively, such that the currents flowing through the PMOS transistors  288  and  290  may be controlled by controlling currents that flow through the NMOS transistors  294  and  296 , respectively. 
   Gates of the NMOS transistors  294  and  296  are coupled to Vap and Vcap voltages  298  and  300 , respectively. The Vap and Vcap voltages  298  and  300  preferably are identical to the Vap and Vcap voltages  254  and  256  provided to the gates of the PMOS transistors  240  and  242 . Thus, as the PMOS transistors  240  and  242  tend to shut off due to increasing Vap and Vcap voltages, the NMOS transistors  294  and  296  tend to open up to conduct increased currents. Therefore, when the Vap and Vcap voltages  254 ,  256  go up to the point where the PMOS transistor  238  (tail current source) enters into the triode region of operation, the differential pair of NMOS transistors  294  and  296  preferably provide sufficient transconductance to prevent abrupt changes to the magnitude of the I 1  and I 2  currents. 
   Sources of the NMOS transistors  294  and  296  are coupled to a drain of an NMOS transistor  302 . A gate of the NMOS transistor  302  is coupled to a bias voltage supplied by a biasing circuit (not shown). The design and use of the biasing circuit to apply suitable potential at the gate of the NMOS transistor  302  is well known to those skilled in the art. A source of the NMOS transistor  302  is coupled to ground. 
   The PMOS transistors  286 ,  288 ,  290 ,  292  and the NMOS transistors  294 ,  296 ,  304  and  306  may be referred to as a differential current mirror circuit, which supplies currents to the transconductance cell  204  as needed. When the Vap and Vcap voltages  254 ,  256  increase, the current through the PMOS transistors  238  decreases. In order to maintain the I 1  and I 2  currents, the currents through the PMOS transistors  286  and  292 , respectively, preferably are supplied to the transconductance cell  204  over the current supply lines  260 ,  258  respectively. In order to channel the currents through the PMOS transistors  286  and  292  to the transconductance cell  204  and not as currents I 3  and I 4  towards ground, NMOS transistors  304  and  306  preferably are used to control the currents I 3  and I 4 . 
   When the transconductance cell is used by itself without a high-swing circuitry, the input swing of the Vap and Vcap voltages have been limited to Vgs of the PMOS transistors  240  and  242  plus the Vdsat of the PMOS transistor  238  (tail current source). With the addition of the high-swing circuitry, the input swing is increased by Vgs of the NMOS transistors  294  and  296  since they supply currents to be the I 1  and I 2  currents when the PMOS transistor  238  enters into the triode region. Since the Vcap voltage is filtered VCO control voltage, the swing of the VCO control voltage is similarly increased. 
   The drains of the PMOS transistors  286  and  292  are coupled to drains of the NMOS transistors  304  and  306 , respectively. Sources of the NMOS transistors  304  and  306  are coupled to ground, respectively. Gates of the NMOS transistors  304 ,  306  are coupled to a gate of an NMOS transistor  284 . A source of the NMOS transistor  284  is coupled to ground, and the gate and a drain of the NMOS transistors  284  are coupled to one another. Thus, the NMOS transistor is configured as a diode, and the NMOS transistors  304  and  306  are current mirrors of the NMOS transistor  284 . 
   As the Vap and Vcap voltages  254  and  256  increase, the I 1  and I 2  currents tend to gradually lose transconductance. In order to make up for the gradual loss to transconductance, the currents supplied over the current supply lines  258  and  260  preferably should increase gradually as well. To this end, the currents I 3  and I 4  preferably should be decreased gradually to provide gradually increasing currents over the current supply lines  258  and  260 . In the embodiment of the present invention depicted in  FIG. 6 , a transistor configuration similar to that of the PMOS transistors  240  and  242  is used to result in the gradual decrease of the I 3  and I 4  currents. 
   The drain of the NMOS transistor  284  is also coupled to drains of PMOS transistors  266  and  268 , which together may be referred to as PMOS input transistors. The first PMOS input transistors are configured similarly to the PMOS transistors  240  and  242  in the transconductance cell  204 . 
   The PMOS transistors  266  and  268  receive Vap and Vcap voltages  270  and  272  at their respective gates. The Vap and Vcap voltages  270  and  272  preferably are provided by the charge pump I/O circuit  202 , and preferably are identical to the Vap and Vcap voltages  254  and  256 . Sources of the PMOS transistors  266  and  268  are coupled to a drain of a PMOS transistor  264  whose source is coupled to the voltage supply. Therefore, as the PMOS transistors  240  and  242  tend to push the PMOS transistor  238  into triode region of operation as the Vap and Vcap voltages  254 ,  256  increase, tending to decrease the I 1  and I 2  currents, the PMOS transistors  266  and  268  tend to push the PMOS transistor  264  into triode region of operation as the Vap and Vcap voltages  270  and  272  increase, tending to decrease the I 3  and I 4  currents. 
   A gate of the PMOS transistor  264  is coupled to a gate of a PMOS transistor  262  whose source is coupled to the voltage supply. The gate and a drain of the PMOS transistor  262  are coupled to one another. Therefore, the PMOS transistor  262  is configured as a current source diode, and the PMOS transistor  264  is configured as a current mirror of the PMOS transistor  262 . 
   When the Vap and Vcap voltages  298  and  300  go down to the point where the NMOS transistor  302  is in triode region of operation, the PMOS transistors  240  and  242  preferably provide sufficient transconductance to supply the I 1  and I 2  currents. However, if the I 3  and I 4  current paths are left unchecked, a portion of currents through the PMOS transistors  240  and  242  may instead be provided as the I 3  and I 4  currents. Therefore, the PMOS transistor  262  should be pushed into triode region of operation as well when the Vap and Vcap voltages are sufficiently low, so that the current flowing through the PMOS transistor  264  and the NMOS transistor  284  are limited, which in turn, limits the I 3  and I 4  currents flowing through the NMOS transistors  304  and  306 , respectively. 
   To this end, the drain of the PMOS transistor  262  is coupled to drains of NMOS transistors  274  and  276 , which may be referred to as first NMOS input transistors. Gates of the NMOS transistors  274  and  276  are coupled to Vap and Vcap voltages  278  and  280 . The Vap and Vcap voltages  278  and  280  preferably are provided by the charge pump I/O circuit  202 , and preferably are identical the Vap and Vcap voltages  254  and  256 . 
   Sources of the NMOS transistors  274  and  276  are coupled to a drain of an NMOS transistor  282  whose source is coupled to ground. A gate of the NMOS transistor  282  is coupled to a bias voltage provided by a biasing circuit (not sown). Design and use of biasing circuits are well known to those skilled in the art. 
   The operation of the high-swing transconductance cell  203  may be described in reference to  FIGS. 7 and 8 .  FIGS. 7 and 8  illustrate a flow diagram of the process of providing currents from the high-swing circuitry  205  to the transconductance cell  204 , such that the current provided to the charge pump I/O circuit  202  is not abruptly decreased when the PMOS transistor  238  (tail current source) enters into the triode region of operation. The  FIGS. 7 and 8  also illustrate the process of recovering from the PMOS transistor  238  (tail current source) being in the triode region. 
   In step  350  of  FIG. 7 , the Vap and Vcap voltages  254  and  256  are relatively low, i.e., not high enough to push the PMOS transistor  238  (tail current source) into triode region of operation. When the Vap and Vcap voltages are not high enough to push the PMOS transistor  238  into the triode region of operation, the I 1  and I 2  currents are provided through the PMOS transistors  240  and  242  in the transconductance cell  204  as indicated in step  352 . 
   When the Vap and Vcap voltages  254  and  256  are substantially low, the Vap and Vcap voltages  298  and  300  are substantially low as well. In this case, as indicated in step  354 , the NMOS transistors  294  and  296  in the differential current mirror circuit tend to shut off. As a result, the currents through the PMOS transistors  288  and  290  tend to be small. Since the PMOS transistors  286  and  292 , which supply the I 3  and I 4  currents, are current mirrors of the PMOS transistors  288  and  290 , respectively, they tend to shut off as well, as indicated in step  356 , and substantially no current is provided to the transconductance cell  204  over the current supply lines  260  and  258  to add to the I 1  and I 2  currents, respectively, as indicated in step  358 . 
   As the output voltage Vc (provided to the VCO to adjust frequency and/or shift phase) increases, the Vap and Vcap voltages increase as well. As the Vap and Vcap voltages increase, the Vgd of the PMOS transistor  238  (tail current source) increases until the PMOS transistor  238  enters into the triode region of operation as indicated in step  360 . The transconductance of the transconductance cell  234  for generating the I 1  and I 2  currents tend to gradually decrease as the PMOS transistors  240  and  242  tend to shut off as indicated in step  362 . 
   While the transconductance of the transconductance cell  204  decreases, the NMOS transistors  294  and  296  in the differential current mirror circuit receive increased Vap and Vcap voltages  298  and  300 , and they tend to turn on as indicated in step  364 . As the NMOS transistors  294  and  296  turn on, gradually increasing currents that flow through the PMOS transistors  288 ,  290 . As indicated in step  366 , the currents flowing through the PMOS transistors  286 ,  292  increase as well since they are current mirrors of the PMOS transistors  288  and  290 , respectively. The process continues to  FIG. 8  as indicated by arrows  368  and  370 . 
   While the PMOS transistors  286  and  292  conduct currents, the PMOS input transistors  266  and  268  tend to shut off in step  372  since the Vap and Vcap voltages  270  and  272  have been increased. Thus, substantially no current flows through the NMOS transistor  284  at sufficiently high Vap and Vcap voltages  270  and  272 . The NMOS transistors  304  and  306  are current mirrors of the NMOS transistor  284 , and therefore, the NMOS transistors  304  and  306  also tend to not conduct the I 3  and I 4  currents as indicated in step  374 . 
   In the absence of substantial currents through the NMOS transistors  304  and  306 , the currents through the PMOS transistors  286  and  292  cannot flow through the NMOS transistors  304  and  306  as the currents I 3  and I 4 , respectively. Therefore, the currents flowing through the PMOS transistors  286  and  292  are provided to the transconductance cell  204  over the current supply lines  260  and  258 , respectively. The currents from the differential current mirror circuit, therefore, supplies currents as indicated in step  376  so that the currents I 1  and I 2  are not substantially reduced when the PMOS transistor  238  is operating in triode region. 
   When it is desired to decrease the VCO frequency and/or to phase shift (towards phase lag) the VCO output, the VCO control signal Vc of the charge pump  201  should be decreased. In step  378 , as the Vc voltage decreases, the Vap and Vcap voltages  254  and  260  decrease as well, and the PMOS transistor  238  may no longer operate in triode region. 
   As the Vap and Vcap voltages  278  and  280  decrease, the NMOS input transistors  274  and  276  tend to shut off as indicated in step  380 , the NMOS transistor  282  in series enters into triode region of operation, and current flowing through the PMOS transistor  262  tends to substantially decrease. Since the PMOS transistor  264  is a current mirror of the PMOS transistor  262 , the PMOS transistor  264  does not let much current flow through it either. Since substantially no current flows through the PMOS transistor  264 , substantially no current flows through the NMOS transistor  284 . 
   Since the NMOS transistors  304  and  306  in series with the currents I 3  and I 4 , respectively, are current mirrors of the NMOS transistor  284 , they tend to shut off as well as indicated in step  382 , and the magnitude of the currents I 3  and I 4  is not substantial. Meanwhile, the NMOS transistors  294  and  296  tend to shut off due to decreased Vap and Vcap voltages  298  and  300 , and the NMOS transistor  302  enters into triode region of operation. In this case, since substantially no current flows through the PMOS transistors  288  and  290 , the current mirror PMOS transistors  286  and  292  preferably conduct substantially no current as well, as indicated in step  384 . 
   Since the Vap and Vcap voltages  254  and  256  have decreased, the PMOS transistors  240  and  242  in the transconductance cell tend to turn on as indicated in step  386 . Since the PMOS transistor  238  is not in the triode region any more, and the PMOS transistors  240  and  242  conduct currents, the I 1  and I 2  currents are generated within the transconductance cell  204 . 
   Although this invention has been described in certain specific embodiments, many additional modifications and variations would be apparent to those skilled in the art. It is therefore to be understood that this invention may be practiced otherwise than as specifically described. Thus, the present embodiments of the invention should be considered in all respects as illustrative and not restrictive, the scope of the invention to be determined by the appended claims and their equivalents. 
   For example, the present invention has been described in reference to PMOS and NMOS transistors. In various different embodiments of the present invention, any other suitable p-channel and n-channel transistors known to those skilled in the art may be used. Further, as those skilled in the art would appreciate, various different semiconductor fabrication technologies, such as, for example, submicron fabrication technologies, may be used during fabrication of devices including the present invention, which may lead to selection of various different voltages as voltage supplies, bias voltages, VCO control voltages, and the like.

Technology Classification (CPC): 7