Patent Abstract:
A complex resonant circuit includes: a first current path performing a first gain control to an AC power signal being supplied; at least one second current path performing a second gain control different from the first gain control to the AC power signal; at least two resonant circuits provided on the respective first and second current paths, having mutually different resonance or antiresonance points for the AC power signals passing through the respective first and second current paths and capturing the respective AC power signals; at least one compensation current path performing a compensation phase shift to the AC power signal; a compensation circuit, provided on the compensation current path, for removing an unnecessary component of the resonant circuit; and an analog operational circuit performing analog addition or subtraction on the AC power signal having passed through the first and second current paths, and the compensation current path.

Full Description:
TECHNICAL FIELD 
       [0001]    The present invention relates to an antiresonant frequency-varying complex resonant circuit which enables a variable antiresonant frequency range to be flexibly set. 
       BACKGROUND ART 
       [0002]    For electronic components which utilize the natural resonant frequency of, e.g., piezoelectric oscillators, a method for connecting reactive elements such as capacitors in parallel is well-known as means for varying the zero phase frequency, i.e., the antiresonant frequency thereof; however, the frequency range itself cannot be varied by changing the physical constants such as of the piezoelectric oscillators. As a result, an attempt to make a wide frequency variable range available would result in degradation in output itself. 
         [0003]    Disclosed in Patent Literature 1 is a circuit for varying the frequency, which gives a relative minimum power at a power summing point, by controlling the ratio of voltages to be applied to a resonant circuit that includes two series resonant circuits. In this circuit, the frequency range with two series resonant frequencies at the respective ends can be arbitrarily controlled by varying the voltage ratio being applied. However, at the center of the variable frequency range, there occurs an extreme deterioration in the effective resonance quality factor Q value which is computed from the frequency range (3 dB bandwidth), in which the effective value of power is twice that at a relative minimum, based on the performance at the relative minimum, that is, the relation between the effective value of power at the relative minimum and the frequency. 
         [0004]    Furthermore, the effective Q values at the ends of the variable frequency range suffer, in practice, significant deterioration when compared with the resonance quality factor Q value without load on the crystal oscillator. 
         [0005]    Means for cancelling the parallel capacitance of the crystal oscillator which restricts the variable frequency range is disclosed in Patent Literature 2; however, the means cannot provide a wide variable frequency range. 
         [0006]    Disclosed in Non-Patent Literature 1 is an approach which allows an oscillator circuit for outputting one fixed frequency to provide an improved effective resonance quality factor Q value as a whole bridge circuit by placing a crystal oscillator on one side of the bridge and selecting arbitrary circuit components on the other sides. However, the frequency cannot be varied over a wide band. 
         [0007]    In summary, conventional complex resonant circuits provided only undesirable performances in practice: the operative resonance quality factor Q value was greatly varied over the entirety of a wide variable frequency range; and significant deterioration was found in the resonance quality factor Q value when compared with the resonance quality factor Q value of the employed resonance element itself. 
       CITATION LIST 
     Patent Literature 
       [0008]    PTL 1: International Publication No. 2006/046672 
         [0009]    PTL 2: Japanese Patent Kokai No. H8-204451 
       Non-Patent Literature 
       [0010]    NPL 1: W. R. Sooy, F. L. Vernon, and J. Munushian: “A Microwave Meacham Bridge Oscillator”, Proc. IRE, Vol. 48, No. 7, pp. 1297-1306, July 1960 
       SUMMARY OF INVENTION 
     Technical Problem 
       [0011]    It is an object of the present invention to provide an antiresonant frequency-varying complex resonant circuit which enables a complex resonant circuit with an oscillator, such as a piezoelectric oscillator, having a good resonance quality factor to achieve a value close to the resonance quality factor Q value with the employed resonance element unloaded and set an antiresonant variable frequency range with a high degree of flexibility over a wide frequency range. 
       Solution to Problem 
       [0012]    To address the aforementioned problems, the antiresonant frequency-varying complex resonant circuit according to the present invention includes: a first current path on which first gain control is provided to an AC power signal being supplied; at least one second current path on which second gain control different in an amount of control from the first gain control is provided to the AC power signal; at least two resonant circuits which are provided each on the respective first and second current paths and which have mutually different resonance points or antiresonance points for the AC power signals passing through the respective first and second current paths and capture the respective AC power signals; at least one compensation current path on which a compensation phase shift is provided to the AC power signal; a compensation circuit, provided on the compensation current path, for removing an unnecessary component of the resonant circuit; and an analog operational circuit for performing analog addition or subtraction on the AC power signal having passed through the first current path, the second current path, and the compensation current path. 
       Advantageous Effects of Invention 
       [0013]    According to the complex resonant circuit of the present invention, a resonance variable frequency range can be set with a high degree of flexibility over a desired variable frequency range without deterioration in effective resonance quality factor Q value. 
     
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         [0014]      FIG. 1  is a circuit diagram illustrating a complex resonant circuit according to a first embodiment of the present invention. 
           [0015]      FIG. 2  is an explanatory view illustrating the effects of the first embodiment of the present invention. 
           [0016]      FIG. 3  is a circuit diagram illustrating a complex resonant circuit according to a second embodiment of the present invention. 
           [0017]      FIG. 4  is a view illustrating an example of frequency characteristics according to a conventional technique. 
           [0018]      FIG. 5  is a view illustrating an example of frequency characteristics for which compensation has been made. 
           [0019]      FIG. 6  is an explanatory view illustrating the uniqueness of a solution to compensation characteristics. 
           [0020]      FIG. 7  is a circuit diagram illustrating a complex resonant circuit according to a third embodiment of the present invention. 
           [0021]      FIG. 8  is a view illustrating the simulation results of exemplary frequency characteristics for which no compensation has been provided. 
           [0022]      FIG. 9  is a view illustrating the simulation results of exemplary frequency characteristics for which compensation has been provided. 
           [0023]      FIG. 10  is an enlarged view illustrating an example of frequency characteristics on the lower end of a variable frequency range. 
           [0024]      FIG. 11  is an enlarged view illustrating an example of frequency characteristics at the center of a variable frequency range. 
       
    
    
     DESCRIPTION OF EMBODIMENTS 
     First Embodiment 
       [0025]      FIG. 1  shows an antiresonant frequency-varying complex resonant circuit according to a first embodiment of the present invention. As shown in  FIG. 1 , the antiresonant frequency-varying complex resonant circuit  1  includes: a reference terminal  2 ; an input terminal  3 ; a first attenuation circuit  9  (Attenuator: ATT 1 ) and a second attenuation circuit (Attenuator: ATT 2 ) for attenuating each the power level of an input signal being supplied at a frequency f from the input terminal  3  through a power distribution circuit  5  and a terminal T 11  or a terminal T 12  into mutually different power levels e 1  and e 2  and then supplying each of the signals at the respective resulting powers to a first resonator circuit  7  or a second resonator circuit  8  via a terminal T 21  or a terminal T 22 , respectively; a first phase shift circuit  11  for providing a phase shift of π+θ1 to the power level of an input signal supplied at the frequency f from the input terminal  3  via the power distribution circuit  5  and a terminal T 13  and then supplying the phase-shifted signal to a first compensation circuit  17  via a terminal T 23 ; the first resonator circuit  7  and the second resonator circuit  8  connected via the terminal T 21  or the terminal T 22  to the first attenuation circuit  9  or the second attenuation circuit  10 , respectively; the first compensation circuit  17  connected via the terminal T 23  to the first phase shift circuit  11 ; a power adder circuit  6  connected via a terminal T 31 , a terminal T 32 , and a terminal T 33  to the first resonator circuit  7 , the second resonator circuit  8 , and the first compensation circuit  17 , respectively; and an output terminal  4  connected to the power adder circuit  6 . Furthermore, the path from the terminal T 11  to the terminal T 31  is defined as a first current path  30 , the path from the terminal T 12  to the terminal T 32  as a second current path  40 , and the path from the terminal T 13  to the terminal T 33  as a first compensation current path  50 . 
         [0026]    Each component of the antiresonant frequency-varying complex resonant circuit  1  shown in  FIG. 1  will be described in more detail. The input terminal  3  of the antiresonant frequency-varying complex resonant circuit  1  of  FIG. 1  is connected to a standard signal generator SG, so that an input signal, which is maintained at a constant output and has a frequency f continuously swept, is applied to the input terminal  3  of the antiresonant frequency-varying complex resonant circuit  1 . The input signal is supplied to each of the first attenuation circuit  9 , the second attenuation circuit  10 , and the first phase shift circuit  11  via the power distribution circuit  5 , and the terminal T 11 , the terminal T 12 , and the terminal T 13 , respectively. 
         [0027]    The first attenuation circuit  9  has an input terminal (not shown), an output terminal (not shown), and an external control terminal CNTR 1 . Control is provided through this external control terminal CNTR 1 , thereby allowing the first attenuation circuit  9  to vary arbitrarily the ratio of the power level at the input terminal and the power level at the output terminal and then output the signal at the resulting power from the output terminal via the terminal T 21  to the first resonator circuit  7 . Note that the input terminal of the first attenuation circuit  9  connects to the terminal T 11 . 
         [0028]    The second attenuation circuit  10  has an input terminal (not shown), an output terminal (not shown), and an external control terminal CNTR 2 . Control is provided through this external control terminal CNTR 2 , thereby allowing the second attenuation circuit  10  to vary arbitrarily the ratio of the power level at the input terminal and the power level at output terminal and then output the signal at the resulting power from the output terminal via the terminal T 22  to the second resonator circuit  8 . Note that the input terminal of the second attenuation circuit  10  connects to the terminal T 12 . 
         [0029]    The first phase shift circuit  11  has an input terminal (not shown) and an output terminal (not shown). The first phase shift circuit  11  provides a phase shift of (π+θ1) to an input signal, which is supplied to the input terminal via the terminal T 13 , and then outputs the phase-shifted signal from the output terminal via the terminal T 23  to the first compensation circuit  17 . 
         [0030]    The first resonator circuit  7  connects to the terminal T 21  and the terminal T 31  and delivers the output therefrom to the output terminal  4  via the terminal T 31  and the power adder circuit  6 . The first resonator circuit  7  includes a parallel circuit which is made up of a series circuit of a coil L 1  and a resistor R 1  disposed between the terminal T 21  and the terminal T 31 , and a capacitor C 1  connected in parallel to the series circuit. 
         [0031]    The second resonator circuit  8  connects to the terminal T 22  and the terminal T 32 , and delivers the output therefrom to the output terminal  4  via the terminal T 32  and the power adder circuit  6 . The second resonator circuit  8  includes a parallel circuit which is made up of a series circuit of a coil L 2  and a resistor R 2  disposed between the terminal T 22  and the terminal T 32 , and a capacitor C 2  connected in parallel to the series circuit. 
         [0032]    The first compensation circuit  17  connects to the terminal T 23 , the terminal T 33 , and the reference terminal  2 , and delivers the output therefrom to the output terminal  4  via the terminal T 33  and the power adder circuit  6 . The first compensation circuit  17  is configured to have a series circuit of a resistor RC 1  and a resistor RC 2  disposed between the terminal T 23  and the terminal T 33 , and a resistor RC 3  disposed between the intermediate point (connection point) of the series circuit and the reference terminal  2 . The first compensation circuit  17  removes a resistance component or an unnecessary component of the first resonator circuit  7  and the second resonator circuit  8 . The input signal applied via such a circuit to the input terminal  3  of the antiresonant frequency-varying complex resonant circuit  1  is supplied to each of the first resonator circuit  7 , the second resonator circuit  8 , and the first compensation circuit  17 . The power level at that time is as follows. That is, The level of power applied to each of the first resonator circuit  7 , the second resonator circuit  8 , and the first compensation circuit  17  is an absolute voltage value of |e 1 |, |e 2 |, and |e 3 |, respectively, in terms of the respective electromotive forces. Here, |e 3 | is the absolute value that is the same as the electromotive force of the standard signal generator SG. This is because on the first compensation current path  50 , a predetermined attenuated power level is not provided. Furthermore, the phase of the first resonator circuit  7  and the second resonator circuit  8  is not shifted (i.e., the phase shift thereof is 0) relative to the input signal applied to the input terminal  3 , whereas only the first compensation circuit  17  is provided with a phase shift of (π+θ1) relative to the input signal applied to the input terminal  3 . Furthermore, at this time, the internal resistance at each of the terminal T 21 , the terminal T 22 , and the terminal T 23  is zs 1 , zs 2 , and zs 3 , respectively. 
         [0033]    That is, the first resonator circuit  7  is equivalent to a series circuit of an equivalent power supply connected to an internal resistor of a resistance value zs 1 , the power supply providing the absolute value of electromotive force |e 1 | with zero phase shift. The second resonator circuit  8  is equivalent to a series circuit of an equivalent power supply connected to an internal resistor of a resistance value zs 2 , the power supply providing the absolute value of electromotive force |e 2 | with zero phase shift. The first compensation circuit  17  is equivalent to a series circuit of an equivalent power supply connected to an internal resistor of a resistance value zs 3 , the power supply providing the absolute value of electromotive force |e 3 | with a phase shift of (π+θ1). 
         [0034]    Now, a description will be made to a modified example of the first embodiment shown in  FIG. 1 . The modified embodiment (not shown) is different from the first embodiment shown in  FIG. 1  in the second current path but the same in the other components. Thus, the descriptions below will be made in relation only to the second current path. 
         [0035]    The second current path of the first embodiment was described to provide gain control to an AC power signal being supplied. The second current path of the modified example relays the AC power signal being supplied. Describing the modified example with reference to  FIG. 1 , the modified example is configured to allow the terminals T 12  and T 22  of  FIG. 1  to be directly connected to each other in place of the second attenuation circuit  10  of  FIG. 1 . Note that in the same manner as in the circuit shown in  FIG. 1 , the modified example also allows for setting a resonance variable frequency range over a desired variable frequency range with a high degree of flexibility without deterioration in effective resonance quality factor Q value. 
         [0036]    Now, a description will be made to the effects and performance of the present invention. Prior to the explanation, the term “Null frequency” will be first defined. It is an object of the present invention to provide an antiresonant frequency-varying complex resonant circuit. The resonance phenomenon of which this complex resonant circuit makes use is not what is called a resonance phenomenon but an antiresonance phenomenon. In general, the characteristics and the performance of the complex resonant circuit can be grasped by examining the operation of the circuit which has a terminal serving as the input terminal thereof and a terminal serving as the output terminal thereof connected between a “high-frequency power supply” and a “load resistance.” 
         [0037]    The complex resonant circuit of the present invention makes use of the antiresonance phenomenon, so that the absolute value of a voltage established across the ends of the aforementioned load resistance exhibits the minimum point. The oscillation frequency at which the absolute value of an output voltage exhibits a relative minimum (also referred to as the minimum point or Null point) will be referred to as the Null frequency and denoted by fnull. The Null frequency is one of those frequencies that characterize the antiresonance phenomenon. 
         [0038]    Now, the effects and performance of the first embodiment will be described in two steps referring to the results of numerical simulations. 
         [0039]    In the first step, it will be described that in a method without the first compensation circuit  17  of the first embodiment, the resonance quality factor Q value at the center of a variable frequency range is significantly deteriorated. In the second step, it will be described that providing a phase shift according to the present invention causes the effective Q value at the center is significantly improved within the entire variable frequency range. 
         [0040]    In summary, the simulation was performed at a center frequency of 10 MHz in a variable frequency range of 1000 ppm (from 9995 kHz to 10005 kHz). For this simulation, the first resonator circuit  7 , the second resonator circuit  8  and the first compensation circuit  17  were given the equivalent circuit constants as shown in Table 1. 
         [0000]    
       
         
               
               
               
             
               
             
           
               
                 TABLE 1 
               
               
                   
               
               
                   
                   
                 FIRST 
               
               
                 FIRST 
                 SECOND 
                 COMPENSATION 
               
               
                 RESONATOR CIRCUIT 
                 RESONATOR CIRCUIT 
                 CIRCUIT 
               
               
                   
               
             
             
               
                 f1 = 9995 kHz 
                 f2 = 10005 kHz 
                 RC1 = 500k Ω 
               
               
                 L1 = 25 mH 
                 L2 = 25 mH 
                 RC2 = 500k Ω 
               
               
                 C1 = 10.142258 fF 
                 C2 = 10.121994 fF 
                 RC3 = 10 Ω 
               
               
                 R1 = 100 Ω 
                 R2 = 100 Ω 
               
               
                 Z S1  = 5 kΩ 
                 Z S2  = 5 kΩ 
                 Z S3  = 5 kΩ 
               
             
          
           
               
                 z 1  = 5 kΩ 
               
               
                   
               
             
          
         
       
     
         [0041]    In  FIG. 2 , the horizontal axis represents the frequency (Hz) and the vertical axis represents the absolute value of a voltage (in volt or V) established across the ends of the load resistance zl.  FIG. 2  illustrates both the results of a simulation performed by a method according to a conventional technique with the first compensation circuit  17  eliminated by applying zero voltage to the first compensation circuit  17  shown in  FIG. 1  and the results of a simulation of the effects of the first embodiment with the first compensation circuit  17  included. 
         [0042]    Since the frequency variation characteristics are substantially symmetric within the variable frequency range,  FIG. 2  shows a curve A and a curve A′ at the low-frequency end thereof and a curve B and a curve B′ at the center. The two curves A′ and B′ correspond to the case with the compensation circuit unavailable, whereas the two curves A and B correspond to the case with the compensation circuit available. 
         [0043]    The two curves A and A′ correspond to the case where the absolute value of a voltage |e 1 | applied to the terminal T 21  and the absolute value of voltage |e 2 | applied to the terminal T 22  are set to 1 V (1 volt) and 0 V (0 volt), respectively. The two curves B and B′ correspond to the case where the absolute value of voltage |e 1 | applied to the terminal T 21  and the absolute value of voltage |e 2 | applied to the terminal T 22  are set to 1 V and 1 V, respectively. Furthermore, the two curves A′ and B′ correspond to the case where the absolute value of voltage |e 3 | applied to the input terminal T 23  of the compensation circuit is set to 0 V with a phase shift of (π+θ1). The two curves A and B correspond to the case where the absolute value of voltage |e 3 | applied to the input terminal T 23  of the compensation circuit is set to 2 1/2  V with a phase shift of (π+θ1). In the simulation, θ1 was set to zero. Accordingly, the phase shift was π. 
         [0044]    A comparison between the relative minimum AS and the relative minimum AS′ showed that the relative minimum AS had dropped more significantly, and likewise, a comparison between the relative minimum BS and relative minimum BS′ showed that the relative minimum BS had dropped more significantly. This means at first glance that the resonance quality factor Q value has been improved. 
         [0045]    That is,  FIG. 2  shows that provision of the first compensation circuit  17  makes it possible to improve the steepness of a drop in the resonance curve on the low-frequency side and at the center of the variable frequency range. To vary the frequency which gives the minimum point of a resonance curve, the ratio of voltages applied to the terminal T 21  and the terminal T 22  is changed; however, it is pointed out that in this embodiment, the voltage applied to the compensation circuit is maintained at a constant absolute value and a constant phase shift. That is, the absolute value and the phase shift thereof need not to be varied or adjusted. This simplifies the circuit structure and provides a high practical value. 
         [0046]      FIG. 2  shows only the low-frequency side of the variable frequency range; however, such an effect can be expected over the entire frequency range. Furthermore, by setting the constants of the compensation circuit and adjusting the absolute value of a voltage applied to the compensation circuit and the phase shift (π+θ1), the resonance quality factor Q value can be so set as to be maintained at a constant value or in a convex or concave shape over the entire frequency range. 
         [0047]    Now, modified examples of the first embodiment will be listed below. The resistor network of the first compensation circuit  17  may be not only a T-type circuit but also a π-type circuit, or alternatively, a series connection of those circuits. Furthermore, the first compensation circuit  17  may be not only a resistor network but also an element including a reactive component. Furthermore, the arm (i.e., the first compensation current path  50 ) with the terminal T 13 , the terminal T 23 , and the terminal T 33 , which are disposed upstream and downstream of the first compensation circuit  17 , may also be provided with an attenuation circuit or an amplifier circuit. 
         [0048]    Now, an example for implementing the resonator circuit using a distributed constant circuit may be an antiresonant frequency-varying complex resonant circuit in which one end of two resonator circuits including strip line paths disposed in close proximity to each of dielectric resonators having mutually different resonance frequencies is connected to a power adder circuit so as to vary the distribution ratio (power ratio) of power to be applied to the other respective terminals of these two strip line paths. 
       Second Embodiment 
       [0049]    A second embodiment is configured such that the resonator circuit thereof has only two piezoelectric oscillators. This configuration has a restriction that a good performance of the resonance quality factor Q value is revealed only in the vicinity of the center of a variable frequency range, but has a feature that the resonator circuit can function in a simplified structure.  FIG. 3  shows an antiresonant frequency-varying complex resonant circuit according to the second embodiment of the present invention. 
         [0050]    The antiresonant frequency-varying complex resonant circuit  100  includes: an input terminal  3 ; a third attenuation circuit  109  for attenuating the power level of an input signal, which is supplied at a frequency f from the input terminal  3  via a power distribution circuit  5 , into a power level e 1  and then supplying the signal at the resulting power to a third resonator circuit  107  via a terminal T 121  and a terminal T 131 ; a fourth attenuation circuit  110  for attenuating the power level of an input signal, which is supplied at the frequency f from the input terminal  3  via the power distribution circuit  5 , into a power level e 2 , and then supplying the signal at the resulting power to a fourth resonator circuit  108  via a terminal T 122  and a terminal T 132 ; a fifth attenuation circuit  113  for attenuating the power level of an input signal, which is supplied at the frequency f from the input terminal  3  via the power distribution circuit  5 , into a power level e 3 , and then supplying the signal at the resulting power to a second phase shift circuit  115  via a terminal T 123 ; and a sixth attenuation circuit  114  for attenuating the power level of an input signal, which is supplied at the frequency f from the input terminal  3  via the power distribution circuit  5 , into a power level e 4 , and then supplying the signal at the resulting power to a third phase shift circuit  116  via a terminal T 124 . Note that power levels e 1 , e 2 , e 3 , and e 4  are different from each other. 
         [0051]    Furthermore, the antiresonant frequency-varying complex resonant circuit  100  includes the second phase shift circuit  115  for providing a phase shift of (π+θ3) to a signal supplied at the frequency f from the fifth attenuation circuit  113  and then supplying the phase-shifted signal to a second compensation circuit  117  via a terminal T 133 ; and the third phase shift circuit  116  for providing a phase shift of (π+θ4) to a signal supplied at the frequency f from the sixth attenuation circuit  114 , and then supplying the phase-shifted signal to a third compensation circuit  118  via a terminal T 134 . Note that the phase shifts (π+θ3) and (π+θ4) are different from each other. 
         [0052]    Furthermore, the antiresonant frequency-varying complex resonant circuit  100  includes the third resonator circuit  107  connected to the third attenuation circuit  109  via the terminal T 121  and the terminal T 131 ; the fourth resonator circuit  108  connected to the fourth attenuation circuit  110  via the terminal T 122  and the terminal T 132 ; the second compensation circuit  117  connected to the second phase shift circuit  115  via the terminal T 133 ; the third compensation circuit  118  connected to the third phase shift circuit  116  via the terminal T 134 ; a power adder circuit  6  connected to each of the terminals T 141 , T 142 , T 143 , and T 144 ; and an output terminal  4  connected to the power adder circuit  6 . 
         [0053]    Each component of the antiresonant frequency-varying complex resonant circuit  100  shown in  FIG. 3  will be described in more detail below. The input terminal  3  of the antiresonant frequency-varying complex resonant circuit  100  of  FIG. 3  is connected to the standard signal generator SG, so that an input signal, which is maintained at a constant output and has a frequency f continuously swept, is applied to the input terminal  3  of the antiresonant frequency-varying complex resonant circuit  100 . 
         [0054]    The input signal applied to the input terminal  3  is supplied to the third attenuation circuit  109 , the fourth attenuation circuit  110 , the fifth attenuation circuit  113 , and the sixth attenuation circuit  114  via the power distribution circuit  5 , and the terminal T 111 , the terminal T 112 , the terminal T 113 , or the terminal T 114 . 
         [0055]    The third attenuation circuit  109  has an input terminal (not shown), an output terminal (not shown), and an external control terminal CNTR 1 . Controlling the external control terminal CNTR 1  would allow the third attenuation circuit  109  to arbitrarily vary the ratio of the power level at the input terminal and the power level at the output terminal and then output the signal at the resulting power from the output terminal via the terminal T 121  and the terminal T 131  to the third resonator circuit  107 . Note that the input terminal of the third attenuation circuit  109  connects to the terminal T 111 . 
         [0056]    The fourth attenuation circuit  110  has an input terminal (not shown), an output terminal (not shown), and an external control terminal CNTR 2 . Controlling the external control terminal CNTR 2  would allow the fourth attenuation circuit  110  to arbitrarily vary the ratio of the power level at the input terminal and the power level at the output terminal and then output the signal at the resulting power from the output terminal via the terminal T 122  and the terminal T 132  to the fourth resonator circuit  108 . Note that the input terminal of the fourth attenuation circuit  110  connects to the terminal T 112 . 
         [0057]    The fifth attenuation circuit  113  has an input terminal (not shown), an output terminal (not shown), and an external control terminal CNTR 3 . Controlling the external control signal CNTR 3  would allow the fifth attenuation circuit  113  to arbitrarily vary the ratio of the power level at the input terminal and the power level at the output terminal and then output the signal at the resulting power from the output terminal via the terminal T 123  to the second phase shift circuit  115 . Note that the input terminal of the fifth attenuation circuit  113  connects to the terminal T 113 . 
         [0058]    The sixth attenuation circuit  114  has an input terminal (not shown), an output terminal (not shown), and an external control terminal CNTR 4 . Controlling the external control signal CNTR 4  would allow the sixth attenuation circuit  114  to arbitrarily vary the ratio of the power level at the input terminal and the power level at the output terminal and then output the signal at the resulting power from the output terminal via the terminal T 124  to the third phase shift circuit  116 . Note that the input terminal of the sixth attenuation circuit  114  connects to the terminal T 114 . 
         [0059]    The second phase shift circuit  115  has an input terminal (not shown) and an output terminal (not shown). The second phase shift circuit  115  provides a phase shift of (π+θ3) to an input signal supplied to the input terminal via the terminal T 123  and then outputs the phase-shifted signal from the output terminal via the terminal T 133  to the second compensation circuit  117 . 
         [0060]    The third phase shift circuit  116  has an input terminal (not shown) and an output terminal (not shown). The third phase shift circuit  116  provides a phase shift of (π+θ4) to an input signal supplied to the input terminal via the terminal T 124  and then outputs the phase-shifted signal from the output terminal via the terminal T 134  to the third compensation circuit  118 . 
         [0061]    The third resonator circuit  107  connects to the terminal T 131  and a terminal T 141  and delivers the output therefrom to the output terminal  4  via the terminal T 141  and the power adder circuit  6 . The third resonator circuit  107  is configured to have a crystal oscillator X 1  disposed between the terminal T 131  and the terminal T 141 . 
         [0062]    The fourth resonator circuit  108  connects to the terminal T 132  and a terminal T 142  and delivers the output therefrom to the output terminal  4  via the terminal T 142  and the power adder circuit  6 . The fourth resonator circuit  108  is configured to have a crystal oscillator X 2  disposed between the terminal T 132  and the terminal T 142 . 
         [0063]    The second compensation circuit  117  connects to the terminal T 133  and terminal T 143  and delivers the output therefrom to the output terminal  4  via a terminal T 143  and the power adder circuit  6 . The second compensation circuit  117  is configured to have a parallel circuit of a capacitor CP 1  and a resistor RP 1  interposed between the terminal T 133  and the terminal T 143 . The second compensation circuit  117  removes an unnecessary component of the third resonator circuit  107 , i.e., a parallel capacitance component CO 1  and a resistance component R 1  of the crystal oscillator X 1 . 
         [0064]    The third compensation circuit  118  connects to the terminal T 134  and the terminal T 144  and delivers the output therefrom to the output terminal  4  via the terminal T 144  and the power adder circuit  6 . The third compensation circuit  118  is configured to have a parallel circuit of a capacitor CP 2  and a resistor RP 2  interposed between the terminal T 134  and the terminal T 144 . The third compensation circuit  118  removes an unnecessary component of the fourth resonator circuit  108 , i.e., a parallel capacitance component CO 2  and a resistance component R 2  of the crystal oscillator X 2 . 
         [0065]    Furthermore, the path from the terminal T 111  to the terminal T 131  is defined as a third current path  130 , the path from the terminal T 112  to the terminal T 132  as a fourth current path  140 , the path from the terminal T 113  to the terminal T 133  as a second compensation current path  150 , and the path from the terminal T 114  to the terminal T 134  as a third compensation current path  160 . 
         [0066]    The input signal applied via such a circuit to the input terminal  3  of the antiresonant frequency-varying complex resonant circuit  100  is supplied to the third resonator circuit  107 , the fourth resonator circuit  108 , the second compensation circuit  117 , and the third compensation circuit  118 . The power level at that time is as follows. 
         [0067]    The level of power applied to each of the third resonator circuit  107  and the fourth resonator circuit  108  is an absolute voltage value of |e 1 | and |e 2 |, respectively, in terms of the respective electromotive forces. The input signal supplied at the frequency f from the input terminal  3  has been provided with a 0 (zero) phase shift for the third resonator circuit  107  and the fourth resonator circuit  108 . Furthermore, at this time, the terminal T 131  and the terminal T 132  have an internal resistance of zs 1  and zs 2 , respectively. 
         [0068]    The level of power applied to each of the second compensation circuit  117  and the third compensation circuit  118  is an absolute voltage value of |e 3 | and |e 4 |, respectively, in terms of the respective electromotive forces. The input signal supplied at the frequency f from the input terminal  3  has been provided with a phase shift of (π+θ3) for the second compensation circuit  117  and a phase shift of (π+θ4) for the third compensation circuit  118 . Furthermore, at this time, the terminal T 133  and the terminal T 134  have an internal resistance of zs 3  and zs 4 , respectively. 
         [0069]    That is, the third resonator circuit  107  is equivalent to a series circuit of an equivalent power supply connected to an internal resistor of a resistance value of zs 1 , the power supply providing the absolute value of electromotive force |e 1 | with zero phase shift. The fourth resonator circuit  108  is equivalent to a series circuit of an equivalent power supply connected to an internal resistor of a resistance value of zs 2 , the power supply providing the absolute value of electromotive force |e 2 | with zero phase shift. The second compensation circuit  117  is equivalent to a series circuit of an equivalent power supply connected to an internal resistor of a resistance value of zs 3 , the power supply providing the absolute value of electromotive force |e 3 | with a phase of (π+θ3). The third compensation circuit  118  is equivalent to a series circuit of an equivalent power supply connected to an internal resistor of a resistance value of zs 4 , the power supply providing the absolute value of electromotive force |e 4 | with a phase of (π+θ4). 
         [0070]    Now, a description will be made to the effects and performance of the second embodiment with reference to the results of numerical simulations. In the first step, it will be described that the effects of parallel capacitance unique to the piezoelectric oscillator can be reduced using the means that can be inferred on the analogy of Patent Literature 2 mentioned above; however, the resonance quality factor Q value of the antiresonant frequency-varying complex resonant circuit  100  will not be increased to such an extent as expected. In the second step, it will be described that the compensation circuit shown in the second embodiment can significantly improve the resonance quality factor Q value. In the third step, it will be described that the resistance value RC 1  and the resistance value RC 2  have an optimum value in a narrow range. 
         [0071]    The outline of the simulation is as follows. The simulation was performed on the assumption that in a variable frequency range of 1000 ppm (from 9995 kHz to 10005 kHz) with a center frequency of 10 MHz, the electromotive forces and the internal resistances of the equivalent power supplies connected equivalently to the terminal T 131  and the terminal T 133  are equal to each other, while the electromotive forces and the internal resistances of the equivalent power supplies connected equivalently to the terminal T 132  and the terminal T 134  are also equal to each other. 
         [0072]    To perform the simulation, the third resonator circuit  107  and the fourth resonator circuit  108  were provided with the equivalent circuit constants shown in Table 2. The second compensation circuit  117  and the third compensation circuit  118  were provided with the equivalent circuit constants shown in Table 3. 
         [0000]    
       
         
               
               
             
               
             
           
               
                 TABLE 2 
               
               
                   
               
               
                 THIRD RESONATOR CIRCUIT 
                 FOURTH RESONATOR CIRCUIT 
               
               
                   
               
             
             
               
                 f1 = 9995 kHz 
                 f2 = 10005 kHz 
               
               
                 L1 = 25.306 mH 
                 L2 = 25.745 mH 
               
               
                 C1 = 10.04976 fF 
                 C2 = 9.799681 fF 
               
               
                 R1 = 10.13 Ω 
                 R2 = 11.311 Ω 
               
               
                 C01 = 3.619 pF 
                 C02 = 3.8237 pF 
               
               
                 Z S1  = 50 Ω 
                 Z S2  = 50 Ω 
               
             
          
           
               
                 z 1  = 50 Ω 
               
               
                   
               
             
          
         
       
     
         [0000]    
       
         
               
               
             
           
               
                 TABLE 3 
               
               
                   
               
               
                 SECOND 
                   
               
               
                 COMPENSATION CIRCUIT 
                 THIRD COMPENSATION CIRCUIT 
               
               
                   
               
             
             
               
                 CP1 = 3.6 pF 
                 CP2 = 3.6 pF 
               
               
                 RP1 = 41 kΩ 
                 RP2 = 41 kΩ 
               
               
                   
               
             
          
         
       
     
         [0073]    A description will be made to the results of the simulation in the first step with reference to  FIG. 4 . For the simulation, the voltages applied to the terminal T 131  to terminal  134  were all equal to 1 V. Furthermore, both the values of part of phase shift θ3 and part of phase shift θ4 were set to 0. Note that in  FIG. 4 , the horizontal axis represents the frequency (Hz) and the vertical axis represents the absolute value of a voltage established across the ends of the load resistance zl. 
         [0074]    Furthermore, the resistors RP 1  and RP 2 , which constitute the second compensation circuit  117  and the third compensation circuit  118  of  FIG. 3 , respectively, were set to infinity, and both the capacitor CP 1  and the capacitor CP 2  were set to 3.6 pF, thereby simulating the means that can be inferred on the analogy of Patent Literature 2 mentioned above. From  FIG. 4 , it will be shown that the influence of a parallel capacitance unique to the piezoelectric oscillator can be alleviated, allowing for exhibiting a single minimum point DS. It will be pointed out that even the conventional technique provides a voltage drop to this extent at the minimum point DS. 
         [0075]    Now, the results of a simulation in the second step will be shown in  FIG. 5 . As shown in  FIG. 5 , the capacitors CP 1  and CP 2  which constitute the second compensation circuit  117  and the third compensation circuit  118  of  FIG. 3 , respectively, were set to the same value, and while the value was being maintained at a constant value of 3.6 pF, the resistors RP 1  and RP 2 , which constitute the second compensation circuit  117  and the third compensation circuit  118 , respectively, were set to the same value and chosen in a wide range in order to perform the simulation. As a result, at 41 kΩ, a sharp drop was found at the relative minimum DS, i.e., a good resonance quality factor Q value was obtained. A comparison between the relative minimum DS of  FIG. 5  and the relative minimum DS of  FIG. 4  shows nearly two orders of magnitude of improvement. As a result, the resonance quality factor Q value at the DS point is significantly improved. At this time, the resonance quality factor Q value reaches 1000000 or six times the resonance quality factor Q value, 150000, of a single crystal oscillator. It can also be found that a parallel circuit of a capacitor of 3.6 pF and a resistor of 41 kΩ employed as the compensation circuit would not have an adverse effect on the characteristics of the series arm of the crystal oscillator. 
         [0076]    Finally, the results of a simulation in the third step are shown in  FIG. 6 . To obtain such an effect as the results of compensation shown in  FIG. 5 , it is necessary to compensate 10Ω, or the value of the series resistor R 1  and the series resistor R 2 , which is a factor that determines the resonance quality factor Q value of the crystal oscillator X 1  of the third resonator circuit  107  and the crystal oscillator X 2  of the fourth resonator circuit  108  in  FIG. 3 . However, according. to the present invention, the second embodiment employed the form of a parallel resistor circuit which simplifies the structure of the compensation circuit, whereby it was found as shown in  FIG. 6  that as the value thereof, an optimum value was present at an unexpected value of a parallel resistance of 41 kΩ. 
         [0077]    In  FIG. 6 , the vertical axis represents the absolute value of a voltage (the value of the relative minimum DS) established across the ends of the load resistance zl of  FIG. 5 , and the horizontal axis represents the value of the resistor RP 1  and the resistor RP 2  which constitute the second compensation circuit  117  and the third compensation circuit  118 , respectively, the resistor RP 1  and the resistor RP 2  being set to an equal value in kΩ and varied as a parameter. It is shown that varying the resistance value on the horizontal axis from 0Ω to infinite Ω results in an optimum point being present only at 41 kΩ. Furthermore, although not illustrated in  FIG. 6 , setting the resistance value on the horizontal axis to less than 1 kΩ results in the absolute voltage value on the vertical axis approaching 1 V. Conversely, setting to above 1000 kΩ would result in the value approaching the value of the minimum point DS (0.001) on the vertical axis of  FIG. 4 , which can be inferred on the analogy of the conventional technique. 
         [0078]    This sole 41 kΩ parallel compensation resistance value is an unexpected value for an equivalent resistance value of 10Ω of the crystal oscillator to be compensated. It is thus noteworthy to have discovered that the resonance quality factor Q value obtained by calculating from the frequency characteristics in the vicinity of the frequency which gives the minimum point DS at this time reaches a value that is six times the resonance quality factor Q value of the crystal oscillator itself being employed. 
         [0079]    Now, as the third resonator circuit  107  and the fourth resonator circuit  108 , a FBAR resonator made of a thin film of aluminum nitride is known to well approximate the resonance characteristics thereof by a parallel circuit which includes a circuit having a series connection of parallel capacitors and a resistor and a series circuit having a series connection of a coil, a capacitor, and a resistor. In such a FRAR resonator, the compensation means of the second embodiment is also effective by appropriately determining the circuit form of the compensation circuit and selecting the circuit constants. 
         [0080]    Now, several more modified items employed will be listed below. The resistor and the capacitor which constitute the compensation circuit may also be connected in series. The third attenuation circuit  109  and the fifth attenuation circuit  113  can be shared with the fourth attenuation circuit  110  and the sixth attenuation circuit  114  so as to half the number of the attenuation circuits. The arm that includes the resonator circuit of  FIG. 3 , for example, the arm of the terminal T 111 , the terminal T 121 , the terminal T 131 , and the terminal T 141  may include a phase shift circuit. 
       Third Embodiment 
       [0081]      FIG. 7  shows an antiresonant frequency-varying complex resonant circuit according to a third embodiment of the present invention. The antiresonant frequency-varying complex resonant circuit  200  of the third embodiment is the antiresonant frequency-varying complex resonant circuit  100  of the second embodiment which further includes two phase shift circuits with the resonator circuit and the compensation circuit modified in structure. Now, a description will be made with reference to  FIG. 7 . 
         [0082]    The antiresonant frequency-varying complex resonant circuit  200  includes a reference terminal  2 ; an input terminal  3 ; a seventh attenuation circuit  209  for attenuating the power level of an input signal, which is supplied at a frequency f from the input terminal  3  via a power distribution circuit  5 , into a power level e 1 , and supplying the signal at the resulting power to a fourth phase shift circuit  211  via a terminal T 221 ; an eighth attenuation circuit  210  for attenuating the power level of an input signal, which is supplied at the frequency f from the input terminal  3  via the power distribution circuit  5 , into a power level e 2 , and supplying the signal at the resulting power to a fifth phase shift circuit  212  via a terminal T 222 ; a ninth attenuation circuit  213  for attenuating the power level of an input signal, which is supplied at the frequency f from the input terminal  3  via the power distribution circuit  5 , into a power level e 3 , and supplying the signal at the resulting power to a sixth phase shift circuit  215  via a terminal T 223 ; and a tenth attenuation circuit  214  for attenuating the power level of an input signal, which is supplied at the frequency f from the input terminal  3  via the power distribution circuit  5 , into a power level e 4 , and supplying the signal at the resulting power to a seventh phase shift circuit  216  via the terminal T 224 . Note that power levels e 1 , e 2 , e 3 , and e 4  are different from each other. 
         [0083]    Furthermore, the antiresonant frequency-varying complex resonant circuit  200  further includes the fourth phase shift circuit  211  for providing a phase shift of θ1 to a signal supplied at the frequency f from the seventh attenuation circuit  209 , and supplying the phase-shifted signal to a fifth resonator circuit  207  via a terminal T 231 ; the fifth phase shift circuit  212  for providing a phase shift of θ2 to a signal supplied at the frequency f from the eighth attenuation circuit  210 , and supplying the phase-shifted signal to a sixth resonator circuit  208  via a terminal T 232 ; the sixth phase shift circuit  215  for providing a phase shift of (θ1+π) to a signal supplied at the frequency f from the ninth attenuation circuit  213 , and supplying the phase-shifted signal to a fourth compensation circuit  217  via a terminal T 233 ; and the seventh phase shift circuit  216  for providing a phase shift of (θ2+π) to a signal supplied at the frequency f from the tenth attenuation circuit  214 , and supplying the phase-shifted signal to a fifth compensation circuit  218  via a terminal T 234 . Note that the phase shifts θ1, θ2, (θ1+π), and (θ2+π) are different from each other. 
         [0084]    Furthermore, the antiresonant frequency-varying complex resonant circuit  200  further includes the fifth resonator circuit  207  connected to the fourth phase shift circuit  211  via the terminal T 231 ; the sixth resonator circuit  208  connected to the fifth phase shift circuit  212  via the terminal T 232 ; the fourth compensation circuit  217  connected to the sixth phase shift circuit  215  via the terminal T 233 ; the fifth compensation circuit  218  connected to the seventh phase shift circuit  216  via the terminal T 234 ; a power adder circuit  6  connected to each of the terminals T 241 , T 242 , T 243 , and T 244 ; and an output terminal  4  connected to the power adder circuit  6 . 
         [0085]    Each component of the antiresonant frequency-varying complex resonant circuit  200  shown in  FIG. 7  will be described in more detail below. The input terminal  3  of the antiresonant frequency-varying complex resonant circuit  200  of  FIG. 7  is connected to the standard signal generator SG, so that an input signal, which is maintained at a constant output and has a frequency f continuously swept, is applied to the input terminal  3  of the antiresonant frequency-varying complex resonant circuit  200 . 
         [0086]    The input signal applied to the input terminal  3  is supplied to the seventh attenuation circuit  209 , the eighth attenuation circuit  210 , the ninth attenuation circuit  213 , and the tenth attenuation circuit  214  via the power distribution circuit  5 , and the terminal T 211 , the terminal T 212 , the terminal T 213 , or the terminal T 214 . 
         [0087]    The seventh attenuation circuit  209  has an input terminal (not shown), an output terminal (not shown), and an external control terminal CNTR 1 . Controlling the external control terminal CNTR 1  would allow the seventh attenuation circuit  209  to arbitrarily vary the ratio of the power level at the input terminal and the power level at the output terminal and then output the signal at the resulting power from the output terminal via the terminal T 221  to the fourth phase shift circuit  211 . Note that the input terminal of the seventh attenuation circuit  209  connects to the terminal T 211 . 
         [0088]    The eighth attenuation circuit  210  has an input terminal (not shown), an output terminal (not shown), and an external control terminal CNTR 2 . Controlling the external control terminal CNTR 2  would allow the eighth attenuation circuit  210  to arbitrarily vary the ratio of the power level at the input terminal and the power level at the output terminal and then output the signal at the resulting power from the output terminal via the terminal T 222  to the fifth phase shift circuit  212 . Note that the input terminal of the eighth attenuation circuit  210  connects to the terminal T 212 . 
         [0089]    The ninth attenuation circuit  213  has an input terminal (not shown), an output terminal (not shown), and an external control terminal CNTR 3 . Controlling the external control signal CNTR 3  would allow the ninth attenuation circuit  213  to arbitrarily vary the ratio of the power level at the input terminal and the power level at the output terminal and then output the signal at the resulting power from the output terminal via the terminal T 223  to the sixth phase shift circuit  215 . Note that the input terminal of the ninth attenuation circuit  213  connects to the terminal T 213 . 
         [0090]    The tenth attenuation circuit  214  has an input terminal (not shown), an output terminal (not shown), and an external control terminal CNTR 4 . Controlling the external control signal CNTR 4  would allow the tenth attenuation circuit  214  to arbitrarily vary the ratio of the power level at the input terminal and the power level at the output terminal and then output the signal at the resulting power from the output terminal via the terminal T 224  to the seventh phase shift circuit  216 . Note that the input terminal of the tenth attenuation circuit  214  connects to the terminal T 214 . 
         [0091]    The fourth phase shift circuit  211  has an input terminal (not shown) and an output terminal (not shown). The fourth phase shift circuit  211  provides a phase shift of θ1 to an input signal supplied to the input terminal via the terminal T 221 , and then outputs the phase-shifted signal from the output terminal via the terminal T 231  to the fifth resonator circuit  207 . 
         [0092]    The fifth phase shift circuit  212  has an input terminal (not shown) and an output terminal (not shown). The fifth phase shift circuit  212  provides a phase shift of θ2 to an input signal supplied to the input terminal via the terminal T 222 , and then outputs the phase-shifted signal from the output terminal via the terminal T 232  to the sixth resonator circuit  208 . 
         [0093]    The sixth phase shift circuit  215  has an input terminal (not shown) and an output terminal (not shown). The sixth phase shift circuit  215  provides a phase shift of (θ1+π) to an input signal supplied to the input terminal via the terminal T 223 , and then outputs the phase-shifted signal from the output terminal via the terminal T 233  to the fourth compensation circuit  217 . 
         [0094]    The seventh phase shift circuit  216  has an input terminal (not shown) and an output terminal (not shown). The seventh phase shift circuit  216  provides a phase shift of (θ2+π) to an input signal supplied to the input terminal via the terminal T 224 , and then outputs the phase-shifted signal from the output terminal via the terminal T 234  to the fifth compensation circuit  218 . 
         [0095]    The fifth resonator circuit  207  connects to the terminal T 231 , the terminal T 241 , and the reference terminal  2 , and delivers the output therefrom via the terminal T 241  and the power adder circuit  6  to the output terminal  4 . The fifth resonator circuit  207  is configured to have a series circuit of a coil LS 1  and a capacitor CS 1  disposed between the terminal T 231  and the terminal T 241 , and the crystal oscillator X 1  disposed between the intermediate point (connection point) of the series circuit and a reference potential  2 . 
         [0096]    The sixth resonator circuit  208  connects to the terminal T 232 , the terminal T 242 , and the reference terminal  2 , and delivers the output therefrom via the terminal T 242  and the power adder circuit  6  to the output terminal  4 . The sixth resonator circuit  208  is configured to have a series circuit of a coil LS 2  and a capacitor CS 2  disposed between the terminal T 232  and the terminal T 242 , and the crystal oscillator X 2  disposed between the intermediate point (connection point) of the series circuit and the reference potential  2 . 
         [0097]    The fourth compensation circuit  217  connects to the terminal T 233 , the terminal T 243 , and the reference terminal  2 , and delivers the output therefrom via the terminal T 243  and the power adder circuit  6  to the output terminal  4 . The fourth compensation circuit  217  is configured to have a series circuit of a coil LS 1 ′ and a capacitor CS 1 ′ disposed between the terminal T 233  and the terminal T 243 , and the resistor RC 1  disposed between the intermediate point (connection point) of the series circuit and the reference potential  2 . The fourth compensation circuit  217  removes an unnecessary component of the fifth resonator circuit  207 , i.e., the resistance component R 1  of the crystal oscillator X 1 . 
         [0098]    The fifth compensation circuit  218  connects to the terminal T 234 , the terminal T 244 , and the reference terminal  2 , and delivers the output therefrom via the terminal T 244  and the power adder circuit  6  to the output terminal  4 . The fifth compensation circuit  218  is configured to have a series circuit of a coil LS 2 ′ and a capacitor CS 2 ′ disposed between the terminal T 234  and the terminal T 244 , and the resistor RC 2  disposed between the intermediate point (connection point) of the series circuit and the reference potential  2 . The fifth compensation circuit  218  removes an unnecessary component of the sixth resonator circuit  208 , i.e., the resistance component R 2  of the crystal oscillator X 2 . 
         [0099]    Furthermore, the path from the terminal T 211  to the terminal T 231  is defined as a fifth current path  230 , the path from the terminal T 212  to the terminal T 232  as a sixth current path  240 , the path from the terminal T 213  to the terminal T 233  as a fourth compensation current path  250 , and the path from the terminal T 214  to the terminal T 234  as a fifth compensation current path  260 . 
         [0100]    The input signal applied to the input terminal  3  of the antiresonant frequency-varying complex resonant circuit  200  via such circuits is supplied to each of the fifth resonator circuit  207 , the sixth resonator circuit  208 , the fourth compensation circuit  217 , and the fifth compensation circuit  218 . The power level at that time is as follows. 
         [0101]    The level of power applied to each of the fifth resonator circuit  207  and the sixth resonator circuit  208  is an absolute voltage value of |e 1 | and |e 2 |, respectively, in terms of the respective electromotive forces. The input signal applied to the input terminal  3  has been provided with a phase shift of θ1 for the fifth resonator circuit  207  and a phase shift of θ2 for the sixth resonator circuit  208 . Furthermore, at this time, the terminal T 231  and the terminal T 232  have an internal resistance of zs 1  and zs 2 , respectively. 
         [0102]    The level of power applied to each of the fourth compensation circuit  217  and the fifth compensation circuit  218  is an absolute voltage value of |e 3 | and |e 4 |, respectively, in terms of the respective electromotive forces. The input signal applied to the input terminal  3  has been provided with a phase shift of (θ1+π) for the fourth compensation circuit  217  and a phase shift of (θ2+π) for the fifth compensation circuit  218 . Furthermore, at this time, the terminal T 233  and the terminal T 234  have an internal resistance of zs 3  and zs 4 , respectively. 
         [0103]    That is, the fifth resonator circuit  207  is equivalent to a series circuit of an equivalent power supply connected to an internal resistor of a resistance value of zs 1 , the power supply providing the absolute value of electromotive force |e 1 | with a phase of θ1. The sixth resonator circuit  208  is equivalent to a series circuit of an equivalent power supply connected to an internal resistor of a resistance value of zs 2 , the power supply providing the absolute value of electromotive force |e 2 | with a phase of θ2. The fourth compensation circuit  217  is equivalent to a series circuit of an equivalent power supply connected to an internal resistor of a resistance value of zs 3 , the power supply providing the absolute value of electromotive force |e 3 | with a phase of (θ1+π). The fifth compensation circuit  218  is equivalent to a series circuit of an equivalent power supply connected to an internal resistor of a resistance value of zs 4 , the power supply providing the absolute value of electromotive force |e 4 | with a phase of (θ2+π). 
         [0104]    Now, the effects and performance of the third embodiment will be explained in three steps referring to the results of numerical simulations. 
         [0105]    In the first step, it will be described that in the method of the third embodiment which does not include the fourth compensation circuit  217  and the fifth compensation circuit  218 , deterioration in the resonance quality factor Q value cannot be ignored at the ends of the variable frequency range. In the second step, it will be described that the compensation circuit of the present invention provides a significant improvement in the resonance quality factor Q value at the ends. In the third step, such a case will be illustrated in which the effective resonance quality factor Q value under an actual operating condition over the entire variable frequency range is so set as to be maintained about at the same value as the resonance quality factor Q value of the single crystal oscillator employed. 
         [0106]    The outline of the simulation is as follows. The simulation was performed on the assumption that in a variable frequency range of 4000 ppm (from 9980 kHz to 10020 kHz) with a center frequency of 10 MHz, the electromotive forces and the internal resistances of the equivalent power supplies connected equivalently to the terminal T 231  and the terminal T 233  are equal to each other, while the electromotive forces and the internal resistances of the equivalent power supplies connected equivalently to the terminal T 232  and the terminal T 234  are equal to each other. 
         [0107]    For the simulation to be performed, the fifth resonator circuit  207  and the sixth resonator circuit  208  were provided with the equivalent circuit constants shown in Table 4. The fourth compensation circuit  217  and the fifth compensation circuit  218  were provided with the equivalent circuit constants shown in Table 5. 
         [0000]    
       
         
               
               
             
               
             
           
               
                 TABLE 4 
               
               
                   
               
               
                 FIFTH RESONATOR CIRCUIT 
                 SIXTH RESONATOR CIRCUIT 
               
               
                   
               
             
             
               
                 f1 = 9980 kHz 
                 f2 = 10020 kHz 
               
               
                 L1 = 25.306 mH 
                 L2 = 25.745 mH 
               
               
                 C1 = 10.04976 fF 
                 C2 = 9.799681 fF 
               
               
                 R1 = 10.13 Ω 
                 R2 = 11.311 Ω 
               
               
                 C01 = 3.619 pF 
                 C02 = 3.8237 pF 
               
               
                 LS1 = 21.392 μH 
                 LS2 = 21.392 μH 
               
               
                 CS1 = 8.200 pF 
                 CS2 = 8.17 pF 
               
               
                 Z S1  = 34.68 Ω 
                 Z S2  = 33.85 Ω 
               
             
          
           
               
                 z 1  = 50 Ω 
               
               
                   
               
             
          
         
       
     
         [0000]    
       
         
               
               
             
           
               
                 TABLE 5 
               
               
                   
               
               
                 FOURTH 
                   
               
               
                 COMPENSATION CIRCUIT 
                 FIFTH COMPENSATION CIRCUIT 
               
               
                   
               
             
             
               
                 f1 = 9980 kHz 
                 f2 = 10020 kHz 
               
               
                 LS1′ = 21.392 μH 
                 LS2′ = 21.392 μH 
               
               
                 CS1′ = 11.819 pF 
                 CS2′ = 11.8407 pF 
               
               
                 RC1 = 2.5 Ω 
                 RC2 = 2.5 Ω 
               
               
                   
               
             
          
         
       
     
         [0108]    Referring to  FIG. 8 , a description will be made to the results of the simulation in the first step. For the simulation, both the voltages applied to the terminal T 233  and the terminal T 234  were 0 V, while as for the phase shifts, the phase shift θ1 provided by the fourth phase shift circuit  211  was +7°, the phase shift θ2 provided by the fifth phase shift circuit  212  was −7°, the phase shift (θ1+π) provided by the sixth phase shift circuit  215  was +187°, and the phase shift (θ2+π) provided by the seventh phase shift circuit  216  was +173°. 
         [0109]    In  FIG. 8 , the horizontal axis represents the frequency (Hz) and the vertical axis represents the absolute value of a voltage established across the ends of the load resistance zl. In this simulation, a numerical experiment was performed by setting an increased amount of attenuation for the ninth attenuation circuit  213  and the tenth attenuation circuit  214  of  FIG. 7 . This caused the fourth compensation circuit  217  and the fifth compensation circuit  218  to be supplied with zero applied voltage and allowed no current to flow into the power adder circuit  6 , thereby preventing the operation of the fourth compensation circuit  217  and the fifth compensation circuit  218  which are to implement the effects of this third embodiment. 
         [0110]    The three curves A, B, and C of  FIG. 8  represent the cases where the voltage e 1  to be applied to the terminal T 231  and the voltage e 2  to be applied to the terminal T 232  were set to 1 V and 0 V, 1 V and 1 V, and 0 V and 1 V, respectively. The three curves have relative minima AS, BS, and CS, respectively, where the two minima, i.e., the relative minimum AS and the relative minimum CS, have a less voltage drop at those relative minima when compared with the relative minimum BS located near the center frequency. At first glance, this shows that the resonance quality factor Q value has deteriorated to such an extent that cannot be ignored. In the next step, the degree of this deterioration is improved by allowing the two compensation circuits of  FIG. 7  to function. 
         [0111]    Now, the simulation of the second step shown in  FIG. 9  was performed by equally setting the electromotive force of the equivalent power supply at both the terminal T 231  and the terminal T 233  of  FIG. 7  and by equally setting the electromotive force of the equivalent power supply at both the terminal T 232  and the terminal T 234 , with the same phase shift as in the case of  FIG. 8 , i.e., with the phase shift θ1 of the fourth phase shift circuit  211  being +7°, the phase shift θ2 of the fifth phase shift circuit  212  being −7°, the phase shift (θ1+π) of the sixth phase shift circuit  215  being +187°, and the phase shift (θ2+π) of the seventh phase shift circuit  216  being +173°. In this case, the two resistors RC 1  and RC 2  of the compensation circuits were set to 10Ω. 
         [0112]    In  FIG. 9 , the horizontal axis represents the frequency (Hz) and the vertical axis represents the absolute value of a voltage established across the ends of the load resistance zl. It can be seen that the degree of voltage drop at both ends is abrupt at the relative minimum AS and the relative minimum CS on the vertical axis when compared with the relative minimum BS at the center. Here, near one of the two relative minima at both ends, for example, in the vicinity of the relative minimum AS, the quotient (i.e., the resonance quality factor Q value) obtained by dividing the frequency, which gives the relative minimum, by the difference between two frequencies (hereinafter referred to as the 3 dB bandwidth) which provides the relative minimum with a value twice as large as the minimum value reaches 1.8 million. This value exceeds 0.15 million, by an order of magnitude, which is the resonance quality factor Q value (i.e., an unloaded Q value) of the single crystal oscillator that constitutes the fifth resonator circuit  207 . This operation can be interpreted such that since the fourth compensation circuit  217  was provided with a setting of 10Ω which is generally the same as the value of the equivalent series resistor R 1  of the crystal oscillator which constitutes the resonator circuit, the loss (resistance) component was cancelled out and thereby substantially nearly completely compensated for at the point of addition of the power adder circuit  6 . Note that for example, the frequency computed from LS 1 ′ and CS 1 ′ shown in Table 5 is consistent with the resonance frequency 9980 kHz of the crystal oscillator. 
         [0113]    The resonance quality factor Q value of 1.8 million (i.e., the effective Q value) was obtained under the operating circuit condition, the value exceeding, by an order of magnitude or greater, the resonance quality factor Q value of 0.15 million of the single crystal oscillator incorporated in the resonator circuit. This phenomenon can be interpreted as follows. That is, according to the present invention, it was found that the resonance characteristic of the Null point is substantially the same as the resonance characteristic of a parallel connection circuit of a coil and a capacitor. Furthermore, since this is a phenomenon at the Null point of a bridge balance, it is conceivably reasonable to exceed the resonance quality factor Q value of the crystal oscillator which constitutes the bridge circuit. 
         [0114]    Finally, the results of a simulation in the third step will be shown in  FIGS. 10 and 11 . In this step, to obtain a constant resonance quality factor Q value across the entire variable frequency range, the values of the shunt resistances of the compensation circuits, i.e., the resistance value RC 1  and the resistance value RC 2  were varied as a parameter for the simulation so as to provide the optimum settings. 
         [0115]    In  FIGS. 10 and 11 , to determine the resonance quality factor Q value, the resonance characteristics in the vicinity of the respective relative minima are shown under magnification, the horizontal axis representing the frequency, the vertical axis representing the absolute value of a voltage established across the ends of the load resistance zl. In  FIG. 10 , to obtain the Null frequency near the lower end of the variable frequency range, the voltage ratios at the terminal T 231  and the terminal T 233  as well as at the terminal T 232  and the terminal T 234  are each 1:0.0625. The resonance characteristics were determined in two ways with the shunt resistance values being 5Ω and 2.5Ω. For 2.5Ω, the resonance quality factor Q value is 130000. This value is generally the same as the resonance quality factor Q value of the single crystal oscillator employed. 
         [0116]    In  FIG. 11 , to obtain the Null frequency near the center of the variable frequency range, the voltage ratios at the terminal T 231  and the terminal T 233  as well as at the terminal T 232  and the terminal T 234  were each set to be 1:1. The resonance characteristics were determined in two ways with the shunt resistance values being 5Ω and 2.5Ω. For 2.5Ω, the resonance quality factor Q value is 150000. This value is generally the same as the resonance quality factor Q value of the single crystal oscillator employed. 
         [0117]    The simulation results thus obtained show that varying the Null frequency over the entire variable frequency range by changing the two applied voltages in a wide range would lead to reduced deterioration in the resonance quality factor Q value across all the frequencies. 
         [0118]    As such, it was possible to adjust the values of RC 1  and RC 2  to thereby keep the resonance quality factor Q value generally constant in the operating condition over the entire variable frequency range. Such resonance quality factor Q values as 130000 and 150000 are generally at the same level as 150000 of the single crystal oscillator, and these numerical values were obtained for the first time by the present invention. 
         [0119]    Now, a description will be made to a modified embodiment. That is, the sixth phase shift circuit  215  may provide a phase shift of (θ1+π) in the combination of a phase shift circuit for providing a phase shift of θ1 and a phase reversal amplifier circuit for providing a phase shift of π or a phase reversal transformer or the like. 
         [0120]    Furthermore, between the input terminal  3  and the output terminal  4 , the order of sequence of the attenuation circuit, the phase shift circuit, and the resonator circuit as well as the order of sequence of the attenuation circuit, the phase shift circuit, and the compensation circuit can be arbitrary, so that the performance of the present invention does not depend on those orders of sequence. The performance of the present invention does not depend on the order of sequence of the coil and the capacitor which constitute the resonator circuit. The phase shift circuit may be implemented by, e.g., a combined circuit of a resistor and a capacitor, a combined circuit of a resistor and an inductive element, a combined circuit of a capacitor and an inductive element, or a delay circuit. Any attenuation circuit may be an amplifier circuit with a variable (gain controllable) amplification factor. When a reversed-phase adder circuit like a differential-input operational amplifier is employed as the power adder circuit, it may be acceptable to employ, as the power distribution circuit, a push-pull-output-like differential-output distribution circuit which has differential-output terminals. The inductive element like a coil may be an element which is equivalently expressed by an active circuit and a resistor. It is possible to widen the variable frequency range by increasing the number of arms including the resonator circuit between the input terminal  3  and the output terminal  4 . The antiresonant frequency-varying complex resonant circuit can be arranged in a series connection, thereby providing an improvement in the quality factor of the frequency selection characteristics of the entire antiresonant frequency-varying complex resonant circuit. 
       REFERENCE SIGNS LIST 
       [0000]    
       
           1  complex resonant circuit 
           2  reference terminal 
           3  input terminal 
           4  output terminal 
           5  power distribution circuit 
           6  power adder circuit 
         SG standard signal generator 
         Z 0  impedance of standard signal generator 
         f frequency outputted by standard signal generator SG 
           7  first resonator circuit 
           8  second resonator circuit 
           9  first attenuation circuit 
           10  second attenuation circuit 
           11  first phase shift circuit 
         zl load resistance 
         CNTR 1 , CNTR 2  control terminal 
           17  first compensation circuit

Technology Classification (CPC): 7