Patent Abstract:
An electronic converter converts high-voltage AC power main voltage, such as 120V, 240V or 277V, to a low-voltage suitable for driving a halogen lamp. The converter includes a rectifier circuit, starter circuit, a driver circuit, a current sensing circuit and a transformer circuit with an optional synchronous output rectifier. The current sensing circuit senses an output current of the converter. The sensed current is used to govern pulse-width modulation of the lamp drive voltage, to provide over-voltage protection. Temperature protection can also be provided to reduce drive current when the converter overheats. This enables reliable operation of the converter over an extended temperature range, and reduces the occurrence of converter component failures due to ground faults or overheating.

Full Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
   The present application is a continuation in part of U.S. patent application Ser. No. 09/899,769 filed Jul. 2, 2001, now U.S. Pat. No. 6,633,139. 

   TECHNICAL FIELD 
   The present invention relates to converters for converting alternating current (AC) power main voltage to a voltage suitable for driving a lamp. 
   BACKGROUND OF THE INVENTION 
   Most electronic converters for converting AC power main voltage to a voltage for driving a lamp, such as a halogen lamp, are based on self-oscillating technology using bipolar transistors. Since bipolar transistors are current operating devices, obtaining feedback for oscillation is relatively simple. However, bipolar transistor converters with or without diode rectification suffer from several disadvantages. For example they are subject to secondary breakdown phenomena, increased current leakage and increased power losses at elevated temperatures. The practical limit for junction temperature is 100° C. (case temperature typically 85° C.). Bipolar transistor converters are also expensive for high voltage applications (for example 277V, 240V and 220V). They also are less efficient in operation than field-effect transistors, because a typical limitation on frequency of operation is 35 kHz due to switching losses. Precise protection against fault conditions is difficult in a simple circuit using bipolar transistors. In addition, size reduction is limited due to operating frequency limitations, and it is difficult to achieve UL Class B temperature classification (130° C. maximum insulation limitation) without a sacrifice in reliability. 
   U.S. Pat. No. 6,157,551 to Barak, et al., assigned to Lightech Electronic Industries Ltd., which issued Dec. 5, 2000, teaches a power converter using bipolar transistors. However, this converter suffers from the foregoing disadvantages. 
   U.S. Pat. No. 6,208,806 to Nerone, assigned to General Electric, which issued Mar. 21, 2001, teaches a power converter using N-channel and P-channel field effect transistors (FETs). Nerone achieves size reduction and improves efficiency by operating at higher frequencies (30 kHz-90 kHz). However, Nerone fails to address the issue of high temperature operation and fault protection. Besides, P-channel FETs are expensive and difficult to obtain compared to N-channel FETs. 
   There therefore exists a need for a converter that is simple and inexpensive to construct, while providing fault protection and achieving reliable, sustained operation at elevated operating temperatures. 
   SUMMARY OF THE INVENTION 
   The present invention provides a converter for converting alternating current (AC) power main voltage to a voltage suitable for driving a lamp. The converter comprises a rectifier circuit connectable to the AC power main, adapted to rectify the AC power main voltage and adapted to provide a direct current (DC) voltage; a driver circuit adapted to receive the unsmoothed DC voltage from the rectifier circuit, and provide a driver output voltage and a driver output current, and further adapted to receive an output current limiting signal; a starter circuit for providing a starter signal that initiates oscillation at an operating frequency in the driver circuit; a sensing circuit for sensing the driver output current and providing the output current limiting signal in response to the sensed driver output current; and a transformer for transforming the driver output voltage to a voltage suitable for driving a lamp such as a halogen lamp. 
   The sensing circuit may be further adapted to provide overheating protection for the converter. Overheating protection can be provisioned in a plurality of ways. In one embodiment, the sensing circuit includes a Negative Temperature Coefficient (NTC) thermistor that is in good thermal contact with the converter. A resistance of the NTC thermistor is reduced as a temperature of the converter rises. This causes the output current limiting signal to reduce output current from the driver circuit when the converter overheats. The reduction in driver output current permits the converter to cool and inhibits component failure. In another embodiment, a silicon diode is used rather than a NTC thermistor. A switching threshold of the silicon diode is reduced as a temperature of the converter rises. This causes the output current limiting signal to output current from the driver circuit to halt the rise in temperature. 
   In accordance with another aspect of the invention, a method is provided for controlling an output voltage of a driver circuit in response to an output current of a converter for converting an AC (alternating current) power main voltage to a voltage suitable for driving a lamp. The method comprises the steps of sensing the converter output current; testing whether the sensed converter output current exceeds a threshold; sensing the extent to which the converter output current exceeds the threshold; triggering a latch when the sensed converter output current exceeds the threshold and stopping an oscillation of the driver circuit; re-setting the latch after a period of time related to an extent to which the converter output current exceeds the threshold, and re-starting the oscillation of the driver circuit. 
   Advantages of the invention include power savings, extended service life for converter components, reduced power loss, and reduced heat generation. 
   A further advantage of the invention is an avoidance of high cost tantalum capacitors, and improved reliability at high temperature operation. 
   Another advantage of the invention is a precise control of output current in addition to protection against fault conditions, such as output short circuits. 
   A further advantage of the invention is an extended operational temperature range for the converter, which enables the converter to achieve an Underwriters Laboratories (UL) Class B temperature classification up to 130° C., which is a maximum insulation limitation. 
   Yet another advantage of the invention is providing a converter with an operating frequency that is greater than 30 kHz, which enables smaller converter packages and more power efficient converters especially when output rectification is MOSFET synchronous. 
   Still another advantage of the invention relates to decreased current leakage and switching losses at elevated temperature resulting from the use of MOSFET (metal oxide silicon field-effect) transistors for switching drive current and rectifying output current. 
   The invention also provides a converter that is reliable, versatile, compact and efficient, with a reduced parts count. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     Further features and advantages of the present invention will become apparent from the following detailed description, taken in combination with the appended drawings, in which: 
       FIG. 1A  is a block diagram of a converter in accordance with the present invention; 
       FIG. 1B  is another block diagram of a converter in accordance with the present invention; 
       FIG. 2  is a schematic diagram of an exemplary rectifier circuit for use in the converter shown in  FIGS. 1A and 1B ; 
       FIG. 3A  is a schematic diagram of an exemplary starter circuit for use in the converter shown in  FIG. 1A ; 
       FIG. 3B  is a schematic diagram of an exemplary starter circuit for use in the converter shown in  FIG. 1B ; 
       FIG. 4A  is a schematic diagram of an exemplary driver circuit for use in the converter shown in  FIG. 1A ; 
       FIG. 4B  is a schematic diagram of an exemplary driver circuit for use in the converter shown in  FIG. 1A ; 
       FIG. 4C  is a schematic diagram of an exemplary driver circuit for use in the converter shown in  FIG. 1B ; 
       FIG. 5A  is a schematic diagram of an exemplary sensing circuit for use in the converter shown in  FIG. 1A ; 
       FIG. 5B  is a schematic diagram of an exemplary sensing circuit for use in the converter shown in  FIG. 1A ; 
       FIG. 5C  is a schematic diagram of an exemplary sensing circuit for use in the converter shown in  FIG. 1A ; 
       FIG. 5D  is a schematic diagram of an exemplary sensing circuit for use in the converter shown in  FIG. 1A ; 
       FIG. 5E  is a schematic diagram of an exemplary sensing circuit for use in&#39;the converter shown in  FIG. 1A ; 
       FIG. 5F  is a schematic diagram of an exemplary sensing circuit for use in the converter shown in  FIG. 1B ; 
       FIG. 6A  is a schematic diagram of an exemplary transformer circuit for use in the converter shown in  FIGS. 1A and 1B ; 
       FIG. 6B  is a schematic diagram of another exemplary transformer circuit for use in the converter shown in  FIGS. 1A and 1B ; 
       FIG. 7  is a plot of an output voltage of the rectifier circuit shown in  FIG. 2 , versus time; 
       FIG. 8  is a plot of an output voltage of the driver circuits shown in  FIGS. 4A ,  4 B, and  4 C, versus time; 
       FIG. 9  is a plot of an output current of the transformer circuit shown in  FIG. 6A , versus time; 
       FIG. 10  is a plot of an output voltage of the transformer circuit shown in  FIG. 6A , versus time; and 
       FIG. 11  is a flowchart of a method of controlling pulse-width modulation in a converter in accordance with the present invention. 
   

   It will be noted that throughout the appended drawings, like features are identified by like reference numerals. 
   DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     FIG. 1A  illustrates a converter  100 A in accordance with the invention. The converter  100  includes a rectifier circuit  104 , a starter circuit  106 A, a driver circuit  108 A, a sensing circuit  110 A, and a transformer circuit  112 A. The rectifier circuit  104  has a first and second input  118 , 120  connectable to an AC (alternating current) power main  102  (shown in dotted outline), a first terminal  122  connected to a power supply node  117  and a second terminal  124  connected to a ground reference node  116 . The starter circuit  106 A has a first terminal  126  connected to power supply node  117 , a second terminal  132  connected to ground reference node  116 , a clamp output  128 , and a starter output  130 . The driver circuit  108 A has a first output  134  connected to the clamp output  128  of starter circuit  106 A, a first input  136  connected to the starter output  130  of starter circuit  106 A, a second input  138 , a first terminal  140  connected to the power supply node  117 , a second output  142  and a second terminal  144 . The sensing circuit  110 A has an output  146  connected to the second input  138  of the driver circuit  108 A, a first terminal  148  connected to the second terminal  144  of the driver circuit  108 A and a second terminal  150  connected to ground reference node  116 . The transformer circuit  112 A has an input  152  connected to the second output  142  of the driver circuit  108 A, a first terminal  154  connected to the power supply node  117 , a second terminal  160  connected to ground reference node  116  and a first and second output  156 , 158  connectable to a lamp  114  (shown in dotted outline). 
     FIG. 1B  illustrates an alternative embodiment of a converter  100 B in accordance with the invention. The converter  100 B shown in  FIG. 1B  is identical to the converter  100 A shown in  FIG. 1A  except that a starter circuit  106 B has a charging output  129  connected to a thermal shutdown terminal  147  of a sensing circuit  10 F. 
     FIG. 2  illustrates a conventional embodiment of the rectifier circuit  104 . The rectifier circuit  104  includes a fuse  202 , an inductor  204 , a resistor  206 , a capacitor  208 , a metal oxide varistor (MOV)  210 , a first diode  212 , a second diode  214 , a third diode  216  and a fourth diode  218 . The fuse  202  is connected between the first input  118  of the rectifier circuit  104  and a first node  220 . Inductor  204  is connected between the first node  220  and a second node  222 . The resistor  206  is connected between the second node  222  and a third node  224 . The capacitor is  208  is connected between the third node  224  and the second input  120  of the rectifier circuit  104 . The MOV  210  is connected between the second node  222  and the second input  120  of the rectifier circuit  104 . The first diode  212  has an anode  226  connected to the second input  120  of the rectifier circuit  104  and a cathode  228  connected to the first terminal  122  of the rectifier circuit  104 . The second diode  214  has an anode  230  connected to the second terminal  124  of the rectifier circuit  104  and a cathode  232  connected to the second input  120  of the rectifier circuit  104 . The third diode  216  has an anode  234  connected to the second node  222  and a cathode  236  connected to the first terminal  122  of the rectifier circuit  104 . The fourth diode  218  has an anode  238  connected to the second terminal  124  of the rectifier circuit  104  and a cathode  240  connected to the second node  222 . 
     FIG. 3A  illustrates a conventional embodiment of the starter circuit  106 A that includes a resistor  302 , a capacitor  305 , a capacitor  306 , a diode  308  and a diac  314 . The resistor  302  is connected between the first terminal  126  of the starter circuit  106 A and a charging node  316 . The capacitor  305  is connected across the resistor  302 , and improves lamp dimming performance in a manner known in the art. The capacitor  306  is connected from the charging node  316  to the second terminal  132  of the starter circuit  106 A. The diode  308  has an anode  310  connected to the charging node  316  and a cathode  312  connected to the clamp output  128  of the starter circuit  106 A. The diac  314  is connected between the charging node  316  and the starter output  130  of the starter circuit  106 A. 
   A starter circuit  106 B of  FIG. 3B  is identical to the starter circuit  106 A shown in  FIG. 3A  except that the charging node  316  is connected to the charging output  129 . 
     FIG. 4A  illustrates a preferred embodiment of the driver circuit  108 A, which includes a high-side switch, preferably a first N-channel FET (field effect transistor)  402 , a low-side switch, preferably a second N-channel FET  410 , a first bi-directional voltage clamping circuit  418 A, a second bi-directional voltage clamping circuit  432 A and a feedback transformer  446 . 
   The first N-channel FET  402  has a gate  404  connected to a first node  472 , a source  406  connected to the first output  134  of the driver circuit  108 A and a drain  408  connected to the first terminal  140  of the driver circuit  108 A. The second N-channel FET  410  has a gate  412  connected to the second input  138 , a source  414  connected to the second terminal  144  of the driver circuit  108 A and a drain  416  connected to the first output  134  of the driver circuit  108 A. 
   The first bi-directional voltage clamping circuit  418 A includes a first zener diode  420  having an anode  422  connected to a second node  474  and a cathode  424  connected to the first node  472 ; and a second zener diode  426 A having an anode  428 A connected to the second node  474  and a cathode  430 A connected to the first output  134  of the driver circuit  108 A. This arrangement of diodes is known as a “back to back” connection. The second bi-directional voltage clamping circuit  432 A includes a third zener diode  434  having an anode  436  connected to a third node  476  and a cathode  438  connected to the second input  138  of the driver circuit  108 A; and a fourth zener diode  440 A having an anode  442 A connected to the third node  476  and a cathode  444 A connected to the second terminal  144  of the driver circuit  108 A. 
   The feedback transformer  446  includes a first winding  448  having a first terminal  450  and a second terminal  452 , a second winding  454  having a first terminal  456  and a second terminal  458 , a third winding  460  having a first terminal  462  and a second terminal  464 , and a fourth winding  466  having a first terminal  468  and a second terminal  470 . The first terminal  450  of the first winding  448  is connected to the second terminal  144  of the driver circuit  108 A. The second terminal  452  of the first winding  448  is connected to the second input  138  of the driver circuit  108 A. The first terminal  456  of the second winding  454  is connected to the ground reference node  116 . The second terminal  458  of the second winding  454  is connected to the first input  136  of the driver circuit  108 A. The first terminal  462  of the third winding  460  is connected to the first node  472 . The second terminal  464  of the third winding  460  is connected to the first output  134  of the driver circuit  108 A. The first terminal  468  of the fourth winding  466  is connected to the first output  134  of the driver circuit  108 A. The second terminal  470  of the fourth winding  466  is connected to the second output  142  of the driver circuit  108 A. 
   The first winding  448 , the second winding  454 , the third winding  460  and the fourth winding  466  of the feedback transformer  446  are arranged so that current flowing into the first terminal  136  of the second winding  454  causes current to flow out of terminal  452  into node  412  and out of node  404  into terminal  462 . 
     FIG. 4B  illustrates an alternative embodiment of the driver circuit  108 B. The embodiment shown in  FIG. 4B  is identical to the embodiment shown in  FIG. 4A  except that the second zener diode  426 A and the fourth zener diode  440 A may be replaced by a first silicon diode  426 B in series with a first resistor  480  and a second silicon diode  440 B in series with a second resistor  484  respectively. Also, the source  414  of the first N-channel FET  410  is connected to the ground reference node  116 ; and the anode  436  of the third zener diode  434  and the anode  442 B of the second silicon diode  440 B are connected to the second terminal  144  of the driver circuit  108 B. 
     FIG. 4C  illustrates another alternative embodiment of the driver circuit  108 C. The embodiment shown in  FIG. 4B  is identical to the embodiment shown in  FIG. 4A  except that the second zener diode  426 A and the fourth zener diode  440 A are in series with the first resistor  480  and the second resistor  484  respectively. Also, the source  414  of the first N-channel FET  410  is connected to the ground reference node  116 ; and the cathode  444 A of the third zener diode  440 A are connected to the second terminal  144  of the driver circuit  108 B. 
     FIG. 5A  illustrates a preferred embodiment of the sensing circuit  110 A, which includes a first resistor  502 , a second resistor  506 , a first diode  508  which is preferably a schottky diode, a first capacitor  514 , a third resistor  516 , a second capacitor  520 , a fourth resistor  522 , an NPN transistor  524 , a PNP transistor  532 , a fifth resistor  540 , a third capacitor  542 , a fourth capacitor  544  and a second diode  546 . 
   The first resistor  502  is connected between the first terminal  148  of the sensing circuit  110 A and the second terminal  150 . of the sensing circuit  110 A. The second resistor  506  is connected between the first terminal  148  of the sensing circuit  110 A and a first node  552 . The first diode  508  has an anode  510  connected to the first node  552  and a cathode  512  that is connected to a second node  554 . The first capacitor  514  is connected between the second node  554  and the second terminal  150  of the sensing circuit  110 A. The third resistor  516  is connected between the second node  554  and a third node  556 . The second capacitor  520  is connected between the third node  556  and the second terminal  150  of the sensing circuit  110 A. The fourth resistor  522  is connected between the third node  556  and the second terminal  150  of the sensing circuit  110 A. The NPN transistor  524  has a base  526  connected to the third node  556 , an emitter  528  connected to the second terminal  150  of the sensing circuit  110 A and a collector  530  connected to a fourth node  558 . The PNP transistor  532  has a base  534  connected to the fourth node  558 , an emitter  536  connected to a fifth node  560  and a collector  538  connected to the third node  556 . The fifth resistor  540  is connected between the fourth node  558  and the fifth node  560 . The third capacitor  542  is connected between the fourth node  558  and the fifth node  560 . The fourth capacitor  544  is connected between the fifth node  560  and the second terminal  150  of the sensing circuit  110 A. The second diode  546  has an anode  548  connected to the output  146  of the sensing circuit  110 A and a cathode  550  connected to the fifth node  560 . For convenience, a portion of sensing circuit  110 A that includes the fourth resistor  522 , the NPN transistor  524 , the PNP transistor  532 , the fifth resistor  540 , the third capacitor  542 , the fourth capacitor  544  and the second diode  546  is hereinafter referred to as a latch  562 . 
     FIG. 5B  illustrates an alternate embodiment of a sensing circuit  110 B. The sensing circuit  110 B is identical to the sensing circuit  110 A except that a negative temperature coefficient (NTC) thermistor  518  has been added in parallel with third resistor  516 . The NTC thermistor  518  provides thermal protection for the converter  100 , as will be explained below in detail. 
     FIG. 5C  shows another alternate embodiment of a sensing circuit  110 C. The sensing circuit  110 C is identical to the sensing circuit  110 A except that the first diode  508  has been replaced with a silicon diode  509  having a cathode  511  connected to the first node  552  and an anode  513  connected to the second node  554 . The silicon diode  509  also provides thermal protection for the converter  100 , as will likewise be explained below in detail. 
     FIG. 5D  shows another alternate embodiment of a sensing circuit  110 D. The sensing circuit  110 D is identical to the sensing circuit  110 B except that the first diode  508  has been replaced with a silicon diode  509  having a cathode  511  connected to first node  552  and an anode  513  connected to second node  554 . Also, the first resistor  502  has been removed. 
     FIG. 5E  shows still another alternate embodiment of a sensing circuit  110 E. The sensing circuit  110 E is identical to the sensing circuit  110 B except that the first resistor  502  has been removed. 
     FIG. 5F  shows yet another alternate embodiment of a sensing circuit  110 F. The sensing circuit  110 E is identical to the sensing circuit  110 D shown in  FIG. 5D  except that: the third resistor  516  and the second capacitor  520  have been removed; a first zener diode  564  having an anode  564 A connected to the third node  556  and a cathode  564 B connected to the second node  554  replaces the third resistor  516 ; the NTC thermistor  518  is connected from the third node  556  to a sixth node  568 ; and a second zener diode  570  has an anode  570 A connected to the sixth node  568  and a cathode  570 B connected to the thermal shutdown terminal  147 . 
     FIG. 6A  shows a conventional embodiment of the transformer circuit  112 A that includes a first capacitor  602 , a second capacitor  604 , and a transformer  606 . The first capacitor  602  is connected between the first terminal  154  of the transformer circuit  112 A and a node  620 . The second capacitor  604  is connected between the node  620  and the second terminal  160  of the transformer circuit  112 A. The transformer  606  has a first winding  608  having a first terminal  610  and a second terminal  612 ; and a second winding  614  having a first terminal  616  and a second terminal  618 . The first terminal  610  of the first winding  608  is connected to the input  152  of the transformer circuit  112 A. The second terminal  612  of the first winding  608  is connected to the node  620 . The first terminal  616  of the second winding  614  is connected to the first output  156  of the transformer circuit  112 A. The second terminal  618  of the second winding  614  is connected to the second output  158  of the transformer circuit  112 A. 
     FIG. 6B  shows an alternative embodiment of the transformer circuit  112 B. The first capacitor  602  is connected between the first terminal  154  of the transformer circuit  112 A and a first node  620 . The second capacitor  604  is connected between the first node  620  and the second terminal  160  of the transformer circuit  112 B. A transformer  630  has: a first winding  632  having a first terminal  632 A connected to the first node  620  and a second terminal  632 B connected to the input  152  of the transformer circuit  112 B; a second winding  634  having a first terminal  634 A connected to the second output  158  of the transformer circuit  112 B and a second terminal  634 B connected to a second node  658 ; a third winding  636  having a first terminal  636 A connected to a third node  666  and a second terminal  636 B connected to the second output  158  of the transformer circuit  112 B; a fourth winding  638  having a first terminal  638 A connected to the second node  658  and a second terminal  638 B connected to a fourth node  646 ; and a fifth winding  640  having a first terminal  640 A connected to a fifth node  652  and a second terminal  640 B connected to the third node  666 . The transformer circuit  112 B also includes: a first N-channel FET  642  having a gate  642 A connected to a sixth node  650 , a source  642 B connected to the seventh node and a drain connected to the first output  156  of the transformer circuit  112 B; a second N-channel FET  644  having a gate  644 A connected to a seventh node, a source  644 B connected to the third node  666  and a drain connected to the first output  156  of the transformer circuit  112 B; a first resistor  648  connected between the fourth node  646  and sixth node  650 ; a second resistor  654  connected between the fifth node  652  and the seventh node  656 ; a third capacitor  660  connected between a the second node  658  and a eighth node  662 ; and a third resistor  664  connected between the eighth node  662  and the third node  666 . 
   The first winding  632 , the second winding  634 , the third winding  636 , the fourth winding  638 , and the fifth winding  640  of the transformer  630  are arranged so that current flowing into the first terminal  632 A of the first winding  632  causes current to flow out of the first terminal  634 A of the second winding  634 , the first terminal  636 A of the third winding  636 , the first terminal  638 A of the fourth winding  638 , and the first terminal  640 A of the fifth winding  640 . 
   In operation, the rectifier circuit  104  ( FIG. 1 ) receives a 60 Hz, 120V power main voltage applied to first and second inputs  118 , 120  and outputs a semi-sinusoidal voltage  702  at 120 Hz, as shown in FIG.  7 . In  FIG. 7 , the x-axis  704  represents time (seconds) and the y-axis  706  represents voltage (Volts). The operation of the rectifier circuit  104  is understood by those skilled in the art. 
   Oscillation of the driver circuit  108 A starts each cycle when the voltage applied to the charging node  316  in the starter circuit  106 A rises sufficiently to turn on the diac  314 . When the diac  314  turns on, a pulse of current is provided to the second winding  454  of the feedback transformer  446 . The pulse of current is coupled through the third winding  460  to the gate  404  of the first N-channel FET  402  and through the second winding  454  to the gate  412  of the second N-channel FET  410 . The direction of the third winding  460  and the second winding  454  are selected so that the pulse of current from the starter circuit  106 A will turn off the first N-channel FET  402  and turn on the second N-channel FET  410 . This causes the voltage on the first output  134  of the driver circuit  108 A to fall. If a load, such as a lamp  114 , is connected to the first and second outputs  156 , 158  of the transformer circuit  112 , then a driver output current will flow through the fourth winding  466 . The direction of the fourth winding  466  is selected so that a positive feedback is supplied to the gate  404  of the first N-channel FET  402  and the gate  412  of the second N-channel FET  410 . The voltage of the first output  134  of the driver circuit  108 A falls to the voltage of the ground reference node  116 . After a period of time determined by the size and a maximum flux density of the core used in the feedback transformer  446 , the feedback to the gate  404  of the first N-channel FET  402  and the gate  412  of the second N-channel FET  410  is removed. The voltage of the first output  134  of the driver circuit  108 A starts to rise, creating a positive feedback that turns on the first N-channel FET  402  and turns off the second N-channel FET  410 . The voltage of the first output  134  of the driver circuit  108 A rises to the voltage of the power supply node  117 . Again, after a period of time determined by the size and the maximum flux density of the core used in feedback transformer  446 , the feedback to the gate  404  of the first N-channel FET  402  and the gate  412  of the second N-channel FET  410  is removed. The voltage of the first output  134  of the driver circuit  108 A then starts to fall, creating positive feedback that turns off the first N-channel FET  404  and turns on the second N-channel FET  410 . Thus, oscillation is established at an operating frequency in the driver circuit  108 . If no load is present, there is no positive feedback and no oscillation occurs. 
   Once oscillation has been established, the diode  312  of the starter circuit  106 A ( FIG. 3 ) maintains a voltage of the charging node  316  of the starter circuit  106 A at a value that is less than a conduction threshold voltage of the diac  314 . 
   Voltage waveform  802  of the first output  134  of the driver circuit  108 A is shown in  FIG. 8 , in which the x-axis  804  represents time (seconds) and the y-axis  806  represents voltage (Volts). The resulting current waveform  902  in the lamp  114  is shown in  FIG. 9 , wherein the x-axis  904  represents time (seconds) and the y-axis  906  represents current (Amperes). It should be noted that the operating frequency illustrated in  FIGS. 8 ,  9 , and  10  is much lower than the normal operating frequency for purposes of clarity, and that normal operating frequency is preferably greater than 43 kHz. 
   The converter  100  provides current overload protection. When a current overload condition occurs, such as a short circuit between the first and second outputs  156 , 158  of transformer circuit  112  causing the output current of driver circuit  108 A to rise above a predetermined threshold, a voltage across the first resistor  502  of the sensing circuit  110 A ( FIG. 5A ) is large enough to turn on the first diode  508  of the sensing circuit  110 A. The first capacitor  514  and the second capacitor  520  are charged so that latch  562  is triggered. The triggering of latch  562  causes current to be drawn into the output  146  of the sensing circuit  110 A and to reduce voltage on the gate  412  of the second N-channel FET  410  and the gate  404  of the first N-channel FET  402  by mutual coupling (FIG.  4 ). This turns off the second N-channel FET  410 , which causes the voltage on the first terminal  148  of the sensing circuit  110 A to decrease, oscillation of the driver circuit  106  then stops, which turns off the first diode  508  of the sensing circuit  110 A. After a period of time determined by values of the first capacitor  514 , the third resistor  516 , the second capacitor  520 , the fourth resistor  522 , the fifth resistor  540 , the third capacitor  542 , the fourth capacitor  544 , and the extent to which the output current of the driver circuit  106  exceeded the predetermined threshold, the latch  562  re-sets to permit oscillation of driver circuit  106  to re-start. The resulting waveform  1002  of the voltage across the lamp  114  is shown in  FIG. 10 , wherein the x-axis  1004  represents time (seconds) and the y-axis  1006  represents voltage (Volts). The voltage across the lamp  114  is thus pulse-width modulated by the current limiting signal on the output  146  of the sensing circuit  110 A. 
   The embodiment shown in  FIG. 5B  introduces the NTC thermistor  518  to provide temperature protection for the converter  100 . The NTC thermistor  518  is placed in good thermal contact with converter  100 . As the temperature of the converter  100  rises, the impedance of the NTC thermistor  518  is reduced. This has the effect of reducing the predetermined threshold for the current overload condition described above. Consequently, as the temperature of the converter  100  increases beyond a threshold determined by resistance characteristics of the NTC thermistor  518 , the driver output current provided to the lamp  114  is reduced, permitting the converter  100  to cool. As cooling occurs, the driver output current is increased. The cycle automatically repeats, as required. 
   In the embodiment shown in  FIG. 5C , the silicon diode  509  serves the same function as the NTC thermistor  518 . The silicon diode  509  is placed in good thermal contact with the converter  100 . As the temperature of the converter  100  rises, the switching threshold of the silicon diode  509  is reduced. This also has the effect of reducing the predetermined threshold of the current limiting circuit described above, to provide thermal protection as described with reference to FIG.  5 B. 
   The embodiment shown in  FIG. 5D  functions substantially the same as the embodiment shown in FIG.  5 B. The removal of the first resistor  502  permits the use of the silicon diode  509  having a higher forward voltage than the schottky diode  508 . 
   The embodiment shown in  FIG. 5E  functions substantially the same as the embodiment shown in FIG.  5 B. 
   In the embodiment shown in  FIG. 5F , the current sensing functions substantially the same as in the embodiment shown in FIG.  5 C. However, the thermal protection functions differently. When the impedance of the thermistor  518  is reduced as the temperature of the converter  100  rises above a predetermined threshold, the latch  562  is triggered by a voltage of the shutdown node  127 . 
   The embodiment of the driver circuit  108 B shown in  FIG. 4B  has the advantage sensing the driver output current indirectly. That is, the driver output current is fed back via the fourth winding  466  of the transformer  468  through the first winding  448  to the second bi-directional voltage clamping circuit  432 . The second resistor  484  of the driver circuit of  FIG. 4B  is used for sensing the driver output current instead of the first resistor  502  of the sensing circuits shown in  FIGS. 5A ,  5 B, and  5 C. 
   The embodiment of the driver circuit  108 A shown in  FIG. 4A  is used in conjunction with the embodiment of the starter circuit  106 A shown in FIG.  3 A and with the embodiments of the sensing circuit  110 A, 110 B, or  110 C shown in  FIGS. 5A ,  5 B and  5 C respectively. The embodiment of the driver circuit  108 B shown in  FIG. 4B  is used in conjunction with the embodiment of the starter circuit  106 A shown in FIG.  3 A and with the embodiments of the sensing circuit  110 D or  110 E shown in  FIGS. 5D and 5E  respectively. The embodiment of the driver circuit  108 C shown in  FIG. 4C  is used in conjunction with the embodiment of the starter circuit  106 B shown in FIG.  3 B and with the embodiment of the sensing circuit  110 F shown in FIG.  5 F. 
   The embodiment of the transformer circuit  112 A shown in  FIG. 6A  is used in conjunction with any of the above combinations of starter circuits  106 A or  106 B, driver circuits  108 A, 108 B or  108 C and sensing circuits  110 A, 110 B, 110 C, 110 D, 110 E or  110 F for providing an AC voltage suitable for driving the lamp  114 . The embodiment of the transformer circuit  112 B shown in  FIG. 6B  is used in conjunction with any of the above combinations of starter circuits  106 A or  106 B, driver circuits  108 A, 108 B or  108 C and sensing circuits  110 A, 110 B, 110 C, 110 D, 110 E or  110 F for providing a DC voltage suitable for driving the lamp  114  wherein the first output  156  is a positive terminal and the second output  158  is a negative terminal. 
   The embodiment shown in  FIG. 6B  functions as a synchronous full-wave rectifier, in which the fourth winding  638  provides a gating voltage to the first FET  642  and the fifth winding  640  provides a gating voltage to the second FET  644 . The third capacitor  660  and second resistor  664  provide filtering of the DC voltage. 
   The invention also provides a method for controlling an output voltage of the driver circuit  106  to provide current limiting protection for the converter  100 .  FIG. 11  is a flowchart  1100  illustrating the method. The method starts (step  1102 ) when power is supplied to the AC inputs  118 , 120  of the rectifier  104 . The driver output current is sensed (step  1104 ) by the sensing circuit  110 A,  110 B,  110 C,  110 D,  110 E, or  110 F to determine whether the sensed driver output current exceeds a threshold (step  1106 ) determined by the component values of the components of the sensing circuit  110 A, as described above. If the driver current is not greater than the threshold, the sensing of the driver output current continues (step  1104 ). If, however, the sensed driver output current exceeds the threshold, then the extent to which the driver output current exceeds the threshold is sensed (step  1108 ). The latch  562  is triggered when the sensed driver output current exceeds the threshold. This stops an oscillation of the driver circuit (step  1110 ). The latch  562  is re-set after a period of time related to an extent to which the driver output current exceeded the threshold (step  1112 ). Meanwhile, the sensing circuit  110 A continues to sense the driver output current (step  1102 ). 
   As explained above, if the NTC thermistor  518  ( FIGS. 5B ,  5 D,  5 E, or  5 F) or the silicon diode  509  ( FIG. 5C ) are added to the sensing circuit  110 , the converter  100  is further provided with temperature protection, which permits the converter  100  to continue to operate at elevated temperatures without component damage. Experimentation has shown that the converter  100  in accordance with the invention can be operated for extended periods of time at case temperatures of at least 110° C., provided that the sensing circuit  110  is constructed as shown in  FIGS. 5B ,  5 C,  5 D,  5 E, or  5 F. 
   The invention therefore provides a simple, high-frequency, light-weight, compact converter  100  that is inexpensive to construct and more robust than converters known from the prior art. The high operating frequency permits all capacitors:  306  shown in  FIGS. 3A and 3B ;  514 , 520 , 542 , 544  shown in  FIGS. 5A-F ;  602 , 604  shown in  FIGS. 6A and 6B ; and  606  shown in  FIG. 6B ; to be solid-state non-polarized capacitors, thereby reducing the weight and package size of the converter  100 . 
   The embodiment(s) of the invention described above is (are) intended to be exemplary only. The scope of the invention is therefore intended to be limited solely by the scope of the appended claims.

Technology Classification (CPC): 7