Patent Abstract:
A method is provided for decimating a digital signal by a factor of M and matching it to a desired channel bandwidth. The method applies the digital signal input samples to a (M−1) stage tapped delay line, downsamples the input samples and the output samples of each tapped delay line stage by a factor of M, and applies each of the M downsampled sample value streams to M allpass IIR filters, respectively. The M allpass IIR filtered sample streams are then summed and scaled by a factor of 1/M. The result can then be filtered by a digital channel filter.

Full Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     The present patent application is a continuation of U.S. patent application Ser. No. 11/611,542, filed on Dec. 15, 2006 now U.S. Pat. No. 8,176,107, which application is related to and claims priority of (a) U.S. Provisional Patent Application, entitled “Multi-Standard Multi-Rate Filter,” Ser. No. 60/751,437, filed 16 Dec. 2005; and (b) U.S. Provisional Patent Application, entitled “Differential Evolution Design Of Polyphase IIR Decimation Filters,” Ser. No. 60/752,619 and filed on 20 Dec. 2005. Each of the foregoing applications is hereby incorporated by reference in its entirety into the present application. 
    
    
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to a high-performance; small die size, and low-power dissipation decimation filter. 
     2. Discussion of the Related Art 
     Conventional analog filter solutions require switching between different filters or different filter components when processing information coded according to different standards or different channel bandwidths. For analog filters to achieve the required narrow transition bandwidths and high stopband attenuations, precise component tolerances are required. Precise component tolerances are difficult to achieve on-chip, necessitating the use of off-chip components, thereby resulting in increased system cost. Additionally, temperature compensation and aging are also often required. 
     Conventional digital filter approaches use finite impulse response (FIR) filters or infinite impulse response (IIR) filters. Conventional FIR filters require large numbers of coefficients to meet the transition band and stopband attenuation requirements. Further, multiple sets of these coefficients are required to support the various coding standards and channel bandwidths. As a result, a large on-chip memory is required. Conventional IIR filters also require many sections to meet such requirements and are sensitive to both coefficient and signal quantization. 
     For a detailed description of the theory and design of FIR digital filters, see Alan Oppenheimer and Ronald Schafer, Digital Signal Processing (Prentice-Hall 1975), especially chapters 5 and 6. Further information regarding conventional filter design may also be found in:
         a Lutovac, M. D. and Milic, L. D., “Design of High-Speed IIR Filters Based on Elliptic Minimal Q-Factors Prototype” (“Lutovac and Milic”), Conf. ETRAN 2002, Banja Vrucica, pp. 103-106 (2002).   b Lutovac, M. D., Tosic, D. V., and Evans, B. L., “Filter Design for Signal Processing—Using MATLAB and Mathematica”, Prentice Hall (2001).   c Harris, F. J., “Multirate Signal Processing—For Communication Systems”, Prentice Hall (2004).   d Krukowski, A. and Kale, I., “DSP System Design—Complexity Reduced IIR Filter Implementation for Practical Applications” (“Krukowski and Kale”), Kluwer Academic Publishers (2003).   e Storn, Rainer, “Designing Nonstandard Filters with Differential Evolution” (“Storn”), IEEE SIGNAL PROCESSING MAGAZINE, January 2005.       

     SUMMARY 
     According to one embodiment of the present invention, a method is provided for decimating a digital signal by a factor of M and matching it to a desired channel bandwidth. The method applies the digital signal input samples to a (M−1) stage tapped delay line, downsamples the input samples and the output samples of each tapped delay line stage by a factor of M, and applies each of the M downsampled sample value streams to M allpass HR filters, respectively. The M allpass IIR filtered sample streams are then summed and scaled by a factor of 1/M. The result can then be filtered by a digital channel filter. 
     The present invention is better understood upon consideration of the detailed description below and the accompanying drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  shows a functional block diagram of a zero-IF/very low-IF (ZIF/VLIF) radio receiver front-end  100 . 
         FIG. 2  is a graph illustrating aspects of the filter requirements in relation to the components of the ZIF/VL1F radio front-end. 
         FIG. 3  is a functional block diagram of IF sampling radio receiver front-end  300 , in accordance with an alternate embodiment of the present invention. 
         FIG. 4  shows an example of an M-path polyphase IIR decimate-by-M filter structure  400 . 
         FIG. 5  shows half-band, 2-path polyphase IIR decimator structure  500  configured in accordance with one embodiment of the present invention. 
         FIG. 6  shows third-band, 3-path polyphase IIR decimator structure  600 , configured in accordance with one embodiment of the present invention. 
         FIG. 7  shows generalized 2-path polyphase IIR filter structure  700 , configured in accordance with one embodiment of the present invention. 
         FIG. 8  shows respective 1-coefficient (“real”) section and a 2-coefficient (“complex”) section IIR allpass filter structures  800  and  850 , respectively, configured in accordance with embodiments of the present invention. 
         FIG. 9  shows 3-coefficient filter  900  which is formed by cascading  3  real sections. 
         FIG. 10  illustrates quad-ratio (2, 3, 4, and 6) decimator multi-standard filter  1000 , configured in accordance with one embodiment of the present invention. 
         FIG. 11  illustrates quad-ratio (2, 3, 4, and 6) multi-standard decimator structure  1100 , configured in accordance with yet a further embodiment of the present invention. 
         FIG. 12  shows an example of the allpass filter sections of H 1  filter  1101  of  FIG. 11 , configured in accordance with the present invention. 
         FIG. 13  shows an example of the allpass filter sections  1106 ,  1107  and  1108  of H 2  filter  1102 , configured in accordance with the present invention. 
         FIG. 14  shows an example of the allpass filter sections  1109 ,  1110  and  1111  of H 3  filter  1102 , configured in accordance with the present invention. 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     In exemplary embodiments, the present invention provides a multi-standard, multi-ratio decimator with improved performance at a lower cost (e.g., smaller die size) and a lower DC power dissipation than was achieved in previously available decimators. In one such embodiment, a quad-ratio (e.g., 2, 3, 4, and 6), multi-standard (e.g., DVB-T/H, ISDB-T and T-DAB) decimator that provides excellent performance with a minimal number of logic gates and low DC power dissipation is provided. 
     The present multi-standard, multi-rate filter trades-off power dissipation with performance. Decreasing the sampling rate reduces power dissipation at the expense of increased aliasing distortion. If the adjacent channel interference (ACI) is small, this added distortion may be acceptable. For example, the sampling rate can be adjusted to a smaller multiple of the orthogonal frequency division multiplexing (OFDM) fast Fourier transform (FFT) sampling rate when the ACI is low, and increased to a higher multiple when the ACI is high. In this way power dissipation is reduced, when possible, without performance degradation. 
     Referring now to  FIG. 1 , a functional block diagram of zero-IF/very low-IF (ZIF/VLIF) radio receiver front-end  100  is shown. Multi-standard, multi-rate filters which are configured in accordance with embodiments of the present invention are provided as the final blocks  101  and  102  of the in-phase (I) and quadrature-phase (Q) paths, respectively. The multi-standard, multirate filters  101  and  102  each have transfer function H(z) and decimation ratio M. As shown in  FIG. 1 , signal F RF  denotes the input RF signal applied to mixers  103  and  104  for the I and Q paths, respectively. Mixers  103  and  104  also receive as inputs the output signal of voltage controlled oscillator (VCO)  105  of frequency F VCO . VCO  105 &#39;s signal received at mixer  104  in the Q path is phase-shifted from that received at mixer  103  in the I path by π/2. The output signals of mixers  103  and  104  are applied as input signals to analog low pass filters (LPF)  107  and  108  each of bandwidth B LP . Thus, the input RF signal is downconverted to zero  1 E, or very-low IF. 
     The output signals of LPFs  107  and  108  are capacitivly coupled to input signals of analog-to-digital (A/D) converters  109  and  110  to remove DC offsets. The A/D converters sample and quantize their input signals at a sampling rate of F S . The resulting sampled and quantized signals are digitally filtered and decimated by multi-standard, multi-rate filters  101  and  102  to provide the respective I(k) and Q(k) output signals. 
       FIG. 2  is a graph illustrating various aspects of the filter requirements in relation to other components of the ZIF/VLIF radio. As shown in  FIG. 2 , frequency F PB  denotes the passband bandwidth, frequency F SB  is the frequency at the high frequency edge of the stop-band, and frequency F S  is the A/D sampling rate, amplitude A PB  is the allowable passband ripple, and amplitude A SB  is the required stopband attenuation. 
       FIG. 3  is a functional block diagram of IF sampling radio receiver front-end  300 , in accordance with an alternate embodiment of the present invention. As shown in  FIG. 3 , multi-standard multi-rate filters  301  and  302  are the final blocks of the in-phase (I) and quadrature-phase (Q) paths, each having transfer function H(z) and decimation ratio M. In this embodiment, an input IF signal of frequency F IF  (e.g., 36 MHz, 36.125 MHz, and 36.17 MHz) is applied to an IF bandpass filter (BPF)  303  of bandwidth B BP . The resulting bandpass signal is capacitivly coupled to A/D converter (of sample rate F s )  304  to remove DC offsets. 
     A/D converter  304  samples and quantizes the bandpass signal of bandpass filter  303 , and provides as input samples to digital mixers  305  and  306  of the I and Q paths. In this case, digital mixers  305  and  306  each also receive as an input digital signal an output digital signal of numerically controlled oscillator (NCO)  307 , which receives a digital word F NCO , representing the NCO frequency. As in IF sampling receiver  100 , NCO  307 &#39;s output signal to the Q path mixer  306  is phase-shifted by π/2 from that of the input signal to mixer  305  of the I path. The sampled and quantized IF samples are thus multiplied by the sine and cosine of NCO  307 &#39;s output phase, respectively. The resulting samples are digitally filtered, and decimated by multi-standard, multi-rate filters  301  and  302  to provide the baseband samples. 
     Digital M-path polyphase infinite impulse response (IIR) filters are suitable for use in decimators with decimation ratio M, for M greater than 1. The M-path polyphase IIR filters are based on an M-tap finite impute response (FIR) filter in which the coefficients are replaced by allpass filters. Polyphase IIR filters provide high stopband attenuation and low passband ripple with a relatively small number of coefficients. 
       FIG. 4  shows an example of an M-path polyphase IIR decimate-by-M filter structure  400 . As shown in  FIG. 4 , input sample x(n) is applied to an M−1 stage, tapped delay line. The input sample and the output signals of the M−1 delay line taps are each downsampled by a factor of M. The M downsampled values are input signals to M allpass filters  401 - 0 ,  401 - 1 , . . . ,  401 -( m− 1). The filter output signals are summed in summer  402  and scaled by 1/M scaler  403  to provide output samples y(m). The transfer function of the M-path polyphase IIR decimator  400  is given by: 
               H   ⁡     (   z   )       =       1   M     ⁢       ∑     k   =   0     M     ⁢       z     -   k       ⁢       A   k     ⁡     (     z   M     )                   
Assuming that each of the M allpass filters  401 - 0 ,  401 - 1 , . . . ,  401 -( m− 1) has N cascaded real sections (i.e. N coefficients), the transfer functions of the allpass filters  401 - 0 ,  401 - 1 , . . . ,  401 -( m− 1) each have the form:
 
                 A   k     ⁡     (   z   )       =       ∏     j   =   1     N     ⁢           ⁢         β     k   ,   j       +     z     -   1           1   +       β     k   ,   j       ⁢     z     -   1                     
Substituting the allpass filter transfer functions into the M-path polyphase IIR decimator  400 &#39;s transfer function then gives:
 
               H   ⁡     (   z   )       =       1   M     ⁢       ∑     k   =   0     M     ⁢       z     -   k       ⁢       ∏     j   =   1     N     ⁢           ⁢         β     k   ,   j       +     z     -   M           1   +       β     k   ,   j       ⁢     z     -   M                           
Thus, the total number of filter coefficients in this example is M×N.
 
     To illustrate some of the polyphase IIR decimators of the present invention,  FIG. 5  shows a half-band, 2-path polyphase IIR decimator structure  500 , configured in accordance with one embodiment of the present invention. As shown in  FIG. 5 , input samples x(n) are decimated by 2 and applied as input samples to allpass filter  501 . Concurrently, the input samples are delayed by one sample, decimated by 2, and applied as input samples to allpass filter  502 . Output signals of filters  501  and  502  are summed in summer  503  and scaled by ½ at scaler  504  to provide the output samples y(n). 
     Similarly,  FIG. 6  shows third-band, 3-path polyphase IIR decimator structure  600 , configured in accordance with one embodiment of the present invention. As shown in  FIG. 6 , input samples x(n) are decimated by 3 and applied to allpass filter  601 . The input samples are also delayed by one sample, decimated by 3, and applied to allpass filter  602 . Further, the input samples are delayed by two samples, decimated by 3, and applied to allpass filter  603 . The filter outputs of allpass filters  601 - 603  are then added together by summer  604  and scaled by ⅓ to form output samples y(n). 
       FIG. 7  shows generalized 2-path polyphase IIR filter structure  700 , configured in accordance with one embodiment of the present invention. As shown in  FIG. 7 , input samples x(n) are applied to allpass filter  701 , and filtered by P filter  703 . The filtered samples are applied as input samples to allpass filter  702 . The filtered output samples of allpass filters  701  and  702  are added together in summer  702  and scaled by ½in scaler  705  to provide output samples y(n). 
     To illustrate the transfer function of the allpass filters of the present invention,  FIG. 8  shows 1-coefficient (“real section”) and a 2-coefficient (“complex section”) IIR allpass filter structures  800  and  850 , respectively, configured in accordance with embodiments of the present invention. As shown in  FIG. 8 , for 1-coefficient filter  800 , the input samples are delayed in element  802  by one sample and added in summer  801  to a scaled difference (by β) between the input samples and samples fed back from the output of IIR allpass filter  800 . The samples fed back from the output of IIR allpass filter  800  are delayed in element  803  by one sample. 
     For 2-coefficient filter  850 , the input samples are delayed two samples, and summed in summer  855  with the output samples of scaling element (by β)  853  and the output samples of scaling element (by α)  854  to provide the filter output samples. The input samples to scaling element  853  are the differences between the input samples of the 2-coefficient filter  850  and the output samples of the filter, delayed by two samples. The input samples to scaling elements  854  are the differences between the output samples of the 2-coefficient filter, delayed one sample, and the input samples of 2-coefficient filter  850 , delayed one sample. 
       FIG. 9  shows 3-coefficient filter  900  which is formed by cascading three real sections. In  FIG. 9 , delay elements (e.g., delay elements  903  and  905 ) are shared between stages. 
     The texts by Lutovac and Milic, and Krukowski and Kale, discussed above, provide detailed descriptions of the theory and design of N-path polyphase IIR filters. These texts disclose algorithms for computing the required allpass filter coefficients. In real world implementations, these filter coefficients are quantized to a finite number of bits. However, quantization by rounding or truncation results in significant filter performance degradation (e.g., larger passband ripple and smaller stopband attenuation). The rounded or truncated coefficients are typically far from optimal for the constrained number of bits. Algorithms for optimizing quantized filter coefficients include “bit-flipping” and “Downhill Simplex Method,” described in Chapter 3 of the Krukowski and Kale text, and “Differential Evolution” (DE) described in the Storn text. 
       FIG. 10  illustrates quad-ratio (2, 3, 4, and 6) decimator multi-standard filter  1000 , configured in accordance with one embodiment of the present invention. As shown in  FIG. 10 , filter  1000  has an input sample rate of F S  and an output sample rate which is selectable between F S /2, F S /3, F S /4, and F S /6, depending on the path selections through the filter structure. 
     For decimation by 2, the input samples bypass filter  1001  (with a transfer function labeled “H 1 ”) and decimation by 2 filter  1005 , pass through filter  1003  (with a transfer function labeled “H 2   D2 ”, with its associated decimation by 2) and filter  1004  (with a transfer function labeled “H 3 ”). For decimation by 3, the input samples pass through filter  1001  (labeled “H 1 ”), bypass decimation by 2 filter  1005 , pass through the filter  1003  (labeled “H 2   D3 ”, with associated decimation by 3), and pass through filter  1004  (labeled “H 3 ”). Similarly, for decimation by 4, the input samples pass through filter  1001 , decimation by 2 filter  1005 , filter  1003  (with its associated decimation by 2), and filter  1004 . For decimate by 6, the input samples pass through filter  1001  and decimation by 2 filter  1004 , filter  1002  (with its associated decimation by 3), and filter  1004 . 
     Table 1 shows examples of filters parameters for a multi-standard, multi-rate filter with component transfer functions H 1 , H 2   D2 , H 2   D3 , H 3   DVB-T , H 3   ISDB-T  and H 3   T-DMB , configured in accordance with the present invention, for supporting DVB-T (also DVB-H, labeled “H 3   DVB-T ”), ISDB-T (labeled “H 3   ISDB-T ”), and T-DAB (also T-DMB, labeled “H 3   T-DMB ”) applications. (The subscript of each transfer function indicates the transfer function&#39;s relative position to the other transfer functions; for example, all the H 2  filters are in parallel relationship with each other, each receiving input samples from the output of the H 1  filter and providing samples to the H 3  filters). In 
     Table 1, frequency F PB  denotes the pass-band bandwidth, frequency F SB  denotes the high-frequency edge of the stop-band, and F S  denotes the A/D sampling rate. Table 2 shows quantized filter coefficients that satisfy the requirements shown in 
     Table 1, expressed as hexadecimal fractions and represent various choices of such coefficients for a multi-standard multi-rate filter configured in accordance with the present invention. In Table 2, A i  denotes the transfer function of the IIR filter in the ith path, and each transfer function Ai may be implemented by multiple coefficients. 
     
       
         
               
               
               
               
             
           
               
                   
                 TABLE 1 
               
               
                   
                   
               
               
                   
                 Filter 
                 F PB /F S   
                 F SB /F S   
               
               
                   
                   
               
             
             
               
                   
                 H 1   
                 0.104 
                 0.385 
               
               
                   
                 H 2   D2   
                 0.208 
                 0.271 
               
               
                   
                 H 2   D3   
                 0.139 
                 0.180 
               
               
                   
                 H 3   DVB-T   
                 0.416 
                 0.459 
               
               
                   
                 H 3   ISDB-T   
                 0.343 
                 0.395 
               
               
                   
                 H 3   T-DAB   
                 0.188 
                 0.229 
               
               
                   
                   
               
             
          
         
       
     
     
       
         
               
               
               
               
               
             
               
               
               
               
               
             
           
               
                   
                 TABLE 2 
               
               
                   
                   
               
               
                   
                   
                 A 0  Coefficients 
                 A 1  Coefficients 
                   
               
               
                   
                 ρ 
                 (β 0i , α 0i ) 
                 (β 1i , α 1i ) 
                 A 2  (β 2i ) 
               
               
                   
                   
               
             
             
               
                   
               
             
          
           
               
                 H 1   
                   
                 44h/200h 
                 127h/200h 
                 — 
               
               
                 H 2   D2   
                 — 
                 8Ch/800h 
                 1EDh/800h 
                 0h/800h 
               
               
                   
                   
                 398h/800h 
                 522h/800h 
                 0h/800h 
               
               
                   
                   
                 667h/800h 
                 77Ah/800h 
                 0h/800h 
               
               
                 H 2   D3   
                 — 
                 49h/800h 
                 F5h/800h 
                 216h/800h 
               
               
                   
                   
                 359h/800h 
                 493h/800h 
                 5A6h/800h 
               
               
                   
                   
                 683h/800h 
                 746h/800h 
                 7Beh/800h 
               
               
                 H 3   DVB-T   
                 579h/800h 
                 431h/800h, B5Eh/800h 
                 538h/800h, C53h/800h 
                 — 
               
               
                   
                   
                 65Eh/800h, D65h/800h 
                 774h/800h, E69h/800h 
               
               
                 H 3   ISDB-T   
                 333h/800h 
                 209h/800h, 6EAh/800h 
                 3C2h/800h, 81Ah/800h 
                 — 
               
               
                   
                   
                 593h/800h, 95Bh/800h 
                 732h/800h, A79h/800h 
               
               
                 H 3   T-DAB   
                 −10Eh/800h 
                 131h/800h, −264h/800h 
                 372h/800h, −2FAh/800h 
               
               
                   
                   
                 595h/800h, −388h/800h 
                 73Eh/800h, −3F6h/800h 
               
               
                   
               
             
          
         
       
     
       FIG. 11  illustrates quad-ratio (2, 3, 4, and 6) multi-standard decimator structure  1100 , configured in accordance with yet a further embodiment the present invention.  FIG. 11  represents replacing each of the H 1 , H 2 , and H 3  filter blocks in  FIG. 10  with the appropriate structure from  FIG. 4 ,  FIG. 5  and  FIG. 6 , with all adder outputs truncated to 16-bits, and all paths scaled to ±1 except for the paths labeled ( 2 ), which are scaled to ±2, and the paths labeled ( 3 ), which are scaled to ±4. 
       FIG. 12  shows an example of the allpass filter sections  1104  and  1105  of H 1  filter  1101  of  FIG. 11 , configured in accordance with the present invention. As discussed above, the adder outputs are truncated to 16-bits and the paths are scaled to ±1 except for the paths labeled ( 2 ), which are scaled to ±2. 
       FIG. 13  shows an example of the allpass filter sections  1106 ,  1107  and  1108  of the H 2  filter  1102  (i.e., H 2   D2  and H 2   D3  filters), configured in accordance with the present invention. As discussed above, the adder outputs are truncated to 16-bits, and the paths are scaled to ±1, except for the inputs and outputs of each of the scaling elements β nm  (n running from 0 to 2 and in running from 1 to 3), which are scaled to ±2. 
       FIG. 14  shows an example of the allpass filter sections  1109 ,  1110  and  1111  of the H 3  filter  1102  (i.e., H 3   DVB-T , H 3   ISDB-T , and H 3   T-DAB  filters), configured in accordance with the present invention. As discussed above, the adder outputs are truncated to 16-bits and most paths are scaled to ±2 (as shown by the associated legends), except for those scaled to ±1 or to ±4, as indicated. 
     Returning now to the zero-IF/very-low IF (ZIF/VLIF) radio receiver front-end  100  shown in  FIG. 1 , receiver front-end  100  processes broadcast digital multimedia signals in VHF and UHF bands. In the broadcasting community, these frequencies are often referred to as Band I, Band II, Band III, Band IV, Band V, and L-Band. 
     Receiver front-end  100  is designed for the DVB-T/H, ISDB-T, and T-DAB broadcast digital multimedia standards, each of which uses an OFDM modulation. The DVB-T/H channels are 5, 6, 7, and 8 MHz. The ISDB-T channels are 6, 7, and 8 MHz. The T-DAB channels are approximately 1.7 MHz. By supporting T-DAB, the receiver front-end also supports the T-DMB standard. 
     Receiver front-end  100  converts RF signals into quantized digital samples with minimal degradation. As shown in  FIG. 1 , the RF signal is mixed with in-phase and quadrature-phase local oscillator signals from VCO  105  where the difference between the RF frequency and the VCO frequency is either zero (ZIF) or very small (VLIF). The output signals of mixers  103  and  1014  are each lowpass filtered by LPFs  107  and  108  to eliminate the higher frequency sum components, leaving the ZIF or VLIF components. 
     To simplify filter implementation, thereby reducing die size (cost) and power dissipation, and minimizing signal distortion, the lowpass bandwidth (B LP ) for LPF  107  and  108  may be selected to be significantly larger than required by the signal bandwidth. For example, in one embodiment, the minimum lowpass bandwidth is one-half of the maximum channel bandwidth (e.g., 4-MHz for an 8-MHz channel). One suitable lowpass filter has a 60-dB bandwidth of 11.5 MHz. The filtered signals are capacitively coupled to A/D converters  109  and  110  to remove any DC offset. A/D converters  109  and  110  sample and quantize the signal at sampling rate F S . The sampling rate should be sufficiently high to prevent aliasing, and be an integer multiple M of the OFDM signals&#39; FFT sampling rates. Approximate sampling rates and integer multiples are shown in Table 3 for an 11.5 MHz B LP . 
     
       
         
               
               
               
               
               
             
               
               
               
               
               
             
           
               
                   
                 TABLE 3 
               
               
                   
                   
               
               
                   
                   
                 Channel Bandwidth 
                   
                   
               
               
                   
                 Standard 
                 (MHz) 
                 M 
                 F S  (Msps) 
               
               
                   
                   
               
             
             
               
                   
               
             
          
           
               
                   
                 DVB-T 
                 8 
                 2 
                 18.3 
               
               
                   
                   
                 7 
                 2 
                 16.0 
               
               
                   
                   
                 6 
                 3 
                 20.5 
               
               
                   
                   
                 5 
                 3 
                 17.1 
               
               
                   
                 ISDB-T 
                 8 
                 2 
                 21.7 
               
               
                   
                   
                 7 
                 2 
                 18.9 
               
               
                   
                   
                 6 
                 2 
                 16.3 
               
               
                   
                 T-DAB 
                 1.707 
                 6 
                 12.3 
               
               
                   
                   
               
             
          
         
       
     
     After the signals are sampled and quantized at A/D converters  109  and  110 , the samples are filtered and decimated by decimators  101  and  102  to obtain samples at the OFDM FFT sampling rate with minimal distortion. Decimators  101  and  102  each have a filter transfer function H(z) and decimate by a factor of M. 
     Quad-ratio (2, 3, 4, and 6), multi-standard (DVB-T/H, ISDB-T, and T-DAB) decimator (QRMSD)  1100  of  FIG. 11  is a suitable filter for implementing each of decimator  101  and  102  of  FIG. 1 . Similarly, QRMSD  1100  is also a suitable filter for implementing QRMSD filters  301  and  302  of  FIG. 3 . 
     As discussed above, decimator  1100  is formed by cascading three filters  1101 ,  1102  and  1103 , having transfer functions H 1 , H 2 , and H 3 , respectively. In some configurations, only one or two of the three filters are required. 
     As shown above, transfer function H 1  can be configured as a half-band, decimate by 2 filter (H 1   D2 ), or as a low-pass filter (LPF) H 1 , by switching in, or out, additional delay elements without changing the coefficients. Transfer function H 2  can be configured as either a half-band, decimate by 2 filter (H 2   D2 ), or as a third-band, decimate by 3, filter (H 2   D3 ), by changing the coefficients. Transfer function H 3  can be configured as any one of a DVB-T LPF, H 3   DVB-T , an ISDB-T LPF, H 3   ISDB-T , or a T-DAB LPF, H 3   T-DAB , by changing the coefficients. 
     The H 1  filter may be designed to meet DVB-T&#39;s decimate-by-4 requirement (i.e., M=4), which are more stringent then those for the ISDB-T and T-DAB standards. Consequently, the same H 1  filter may be used for all three standards. Further, an H i  filter designed for M=4 can also be used for M=6 and M=3, so that the same H 1  filter may be used for all standards requiring decimations by 3, 4, and 6. 
     The H 2  filter may be designed to meet the DVB-T requirements, which are again more stringent then those for the ISDB-T and T-DAB standards. In that manner, the same H 2  filter may be used for all three standards. The H 2   D2  filter is substantially identical for M=4 and M=2, and the H 2   D3  filter is substantially identical for M=6 and M=3. Thus, only one H 2   D2  and one H 2   D3  filter are required for all three standards and all decimation ratios 2, 3, 4, and 6. 
     The H3 stop-band rejection filters are substantially identical for all decimation ratios. 
     As mentioned above, the filter requirements are shown in 
     Table 1. 
     As shown in Table 2, single sets of H 1 , H 2   D2 , and H 2   D3  filter coefficients, and three sets of H 3  filter coefficients are required to support all three standards and all four decimation ratios. 
     Thus, a multi-standard, multi-ratio decimator has been described. 
     The detailed description above is provided to illustrate specific embodiments of the present invention and is not intended to be limiting. Many modifications and variations within the scope of the present invention are possible. The present invention is set forth in the following claims.

Technology Classification (CPC): 7