Patent Abstract:
An electronic signal processor for processing signals includes a complex first filter, one or more gain stages and a second filter. The first filter is characterized by a frequency response curve that includes multiple corner frequencies, with some corner frequencies being user selectable. The first filter also has at least two user-preset gain levels which may be alternately selected by a switch. Lower frequency signals are processed by the first filter with at least 12 db/octave slope, and preferably with 18 db/octave slope to minimize intermodulation distortion products by subsequent amplification in the gain stages. A second filter provides further filtering and amplitude control. The signal processor is particularly suited for processing audio frequency signals. Related methods include filtering the input signal with an input filter of the second or third order high pass type, amplifying the filtered signal and further filtering the amplified signal with a low pass filter, which may be of the second order type.

Full Description:
RELATED APPLICATIONS 
     This application is a continuation of application Ser. No. 10/623,433 filed Jul. 18, 2003 now U.S. Pat. No. 7,390,960, entitled Electronic Signal Processor, which is herein incorporated by reference in its entirety. 
    
    
     FIELD OF THE INVENTION 
     The present invention relates generally to electronic signal processors. More particularly, a preferred embodiment of the invention relates to altering or controlling the tonal qualities of electronic signals, such as audio signals, and related methods. 
     BACKGROUND OF THE INVENTION 
     Various prior art devices exist for modifying the tonal qualities of electronic signals. In audio frequency applications, the types of signals processed can be speech, musical instruments, synthesized waveforms, and the like. Prior art devices for processing musical instrument signals generally have a very limited ability to provide the musician with a variety of tonal qualities in the resulting sound. For example, prior art circuits exist for processing electric guitar signals that have a singular tonal quality, or “sound”. This is a serious limitation, since the guitarist must frequently employ a plurality of different circuits if different “sounds” are desired. 
     Some schemes exist in the art that include circuits with more than a singular sound. Usually this involves adding additional active circuits that the guitarist can activate, as desired. While such an arrangement can be successful, it also results in much greater total component count and added expense. 
     In addition, in some applications, it is desirable to deliberately add distortion to the sound to affect the tonal qualities. For example, deliberately adding distortion to the sound of an electric guitar began in the 1950&#39;s when rock music was becoming popular. At this time, the only techniques that an electric guitarist has to increase the amount of distortion into his sound was to increase the volume of a vacuum tube amplifier by (1) picking the strings of the guitar harder, (2) turning the volume of the guitar higher, or (3) turning the volume of the amplifier up; or some combination or variation of all three techniques. However, these techniques have the drawbacks that the guitarist usually could still not achieve the desired level of distortion, and/or high sound pressure levels were created that many people find uncomfortable or even distressing. 
     During the 1960&#39;s, the characteristic sound of an overdriven vacuum tube amplifier was realized while playing at lower volumes by using new types of circuits. These new circuits were frequently called “fuzzboxes” and were separate boxes that were external to the amplifier. Fuzzboxes typically employed a cascade or series connection of two or more transistor amplifier gain stages that had high input-to-output gain and that were easily overdriven by the output signal from the guitar. This provided a favorable increase in distortion and sustain to the guitar sound. However, it also introduced a new quality to the sound that is disliked by many guitarists. This quality is often referred to as the “solid-state sound” or the “transistor sound”. Either of these terms has acquired a very negative connotation to many guitarists. That is, the solid-state or transistor sound is quite different than the “tube sound”, which was developed by the overdriven vacuum tube amplifiers. 
     Many guitarists continue to believe that the best distortion sounds come from amplifiers that employ tube circuits. While the best solid-state amplifiers come close, they are frequently considered to be inferior to the tube amplifiers. Despite the many solid-state amplifiers that have been developed and introduced to the marketplace since the 1960&#39;s, the solid-state sound is still not on par with that of the tube amplifiers. Indeed, many different schools of thought exist on why there are differences in the sound and feel between the solid-state and tube amplifiers. Recent attempts to emulate the sound and feel of tube amplifiers have stagnated. 
     It has been an objective in the guitar industry for many years to develop solid-state amplifiers that have the sound and feel of the overdriven tube amplifier. “Feel” indicates that a tube amplifier also has a certain tactile quality when overdriven. Many guitarists think that the tube amplifiers respond to the guitarists “touch”, including their picking techniques and playing style, better than the solid-state amplifiers. In this respect, it is frequently stated that tube amplifiers are very touch sensitive. 
     There has been a long-felt need for a solid-state amplifier or signal processor that emulates the sound and feel of an overdriven vacuum tube amplifier. 
     A need also exists for a signal processor that emulates the sound of an overdriven vacuum tube amplifier in which the tone may be adjusted or customized to the user&#39;s desires. 
     Accordingly, it is a general object of the present invention to provide a new and improved signal processor that emulates the sound and feel of an overdriven vacuum tube amplifier. 
     Another object of the present invention is to provide a signal processor of the solid-state type that emulates the desired performance characteristics of a tube amplifier. 
     Yet another object of the present invention is to provide a signal processor with sound characteristics that may be adjusted to the user&#39;s tastes. 
     A further object of the present invention is to filter the lower frequency input signals with a second order or third order high pass filter before amplification of the input signals to reduce lower frequency intermodulation distortion when the amplifier is overdriven. 
     A still further object of the present invention is to provide at least two individual gain controls with overlapping gain characteristics that may be switched to provide selectable gain of those frequencies in the passband of the input filter. 
     Another object of the present invention is to provide related methods of filtering an input signal with an input filter of the second or third order high pass type to substantially reduce lower frequency intermodulation distortion in the signal processor. 
     BRIEF SUMMARY OF THE INVENTION 
     This invention is directed to an electronic signal processor that has improved ability to alter the tonal characteristics of an audio frequency input signal and to reduce lower frequency intermodulation distortion. The signal processor may have a buffer stage to receive the input signal and to provide an input signal with low output impedance to the first filter of the signal processor. 
     A first filter is preferably a second or third order high pass filter with a frequency response curve of 12 db/octave slope or 18 db/octave slope for the lower frequencies, respectively. One of the purposes of the first filter is to substantially reduce lower frequency intermodulation distortion by means of such filtering. The first filter also has at least some user-selectable corner frequencies in its frequency response curve so that the user may customize the tonal quality of the signal processor. The first filter preferably also includes at least two adjustable gain levels with overlapping gain characteristics that may be pre-set by the user and that may be alternately selected. The multiple, user-preset, selectable gain levels allow the user to adjust the amount of distortion present in, and therefore the tonal color of, the processor output. 
     The output of the first filter is input to one or more limiting gain stages, which are in series or cascade configuration. These gain stages can increase the amount of distortion present in the processor output. Oppositely poled diodes in the feedback circuits of the amplifiers in the gain stages limit the output amplitude of the amplifiers and contribute to the distortion characteristics of the signal processor. Preferably, the gain stages have an additional or second feedback circuit that introduces a controlled amount of hysteresis, a nonlinear distortion, in the amplification characteristic of the gain stages. Thus, when the gain stages are overdriven by the input signal, the clipping or distortion in the output signal of the gain stages will be enhanced. 
     The present invention also relates to amplifiers with two feedback loops for use in the gain stages of signal processors. The first feedback loop includes a resistor, a capacitor and at least two diodes, with the diodes oppositely poled between the output of the amplifier and its inverting input. The second feedback circuit includes at least one resistor and at least one capacitor coupled between the output of the amplifier and the input of the gain stage. A resistor preferably couples the second feedback loop to the inverting input of the amplifier. The two feedback loops interact to enhance the distortion when the amplifier is overdriven by an input signal. 
     The output from the gain stages is input to a second filter, which is of the low pass type and preferably of the second order low pass type. The output the second filter is provided as the output of the signal processor. 
     Related methods of processing an input signal that includes a band of frequencies to reduce lower frequency intermodulation distortion includes filtering the input signal with the first filter of the second or third order type, supplying the filtered signal to the gain stages, amplifying the filtered signal in the gain stages, supplying the amplified signal to a second filter of the low pass type, filtering the amplified signal in the second filter, and supplying the signal from the second filter as the output signal of the signal processor. The methods also include changing at least some of the corner frequencies in the frequency response curve of the first filter to change or customize the frequency response of the first filter. The methods further include selecting one of the two gain controls in the first filter. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The features of the present invention which are believed to be novel are set forth with particularity in the appended claims. The invention, together with the further objects and advantages thereof, may best be understood by reference to the following description taken in conjunction with the accompanying drawings, in the several figures in which like reference numerals identify like elements, and in which: 
         FIG. 1  is a block diagram of the signal processor of the present invention; 
         FIG. 2  is a schematic circuit diagram of a preferred embodiment of the signal processor of the present invention; 
         FIG. 3  is a schematic circuit diagram of a preferred embodiment of an input filter for the signal processor shown in  FIGS. 1 and 2 ; 
         FIG. 4  is a frequency response curve of the input filter of the block diagram shown in the schematic circuit diagram of  FIG. 3  under selected circuit conditions; 
         FIG. 5  is a frequency response curve of the input filter shown in the schematic circuit diagram of  FIG. 3  under selected circuit conditions; 
         FIG. 6  is a frequency response curve of the input filter shown in the schematic circuit diagram of  FIG. 3  under selected circuit conditions; 
         FIG. 7  is a frequency response curve of the input filter shown in the schematic circuit diagram of  FIG. 3  under selected circuit conditions; 
         FIG. 8  is a schematic circuit diagram of a preferred embodiment of an amplifier stage for the signal processor shown in  FIG. 2 ; 
         FIG. 9  is a schematic circuit diagram of an alternate embodiment of an amplifier stage for the signal processor shown in  FIG. 2 ; 
         FIG. 10  is a frequency response curve of the amplifier stages shown in the schematic circuit diagrams of  FIGS. 8 and 9 ; 
         FIG. 11  is a schematic circuit diagram of an output filter for the signal processor shown in  FIG. 1 ; 
         FIG. 12  is a frequency response curve of the output filter shown in the schematic circuit diagram of  FIG. 11  under selected circuit conditions; 
         FIG. 13  is a block diagram that is related to the block diagram of  FIG. 1 , but with the preferred frequency responses of the first and second filters inserted in the respective filter blocks; 
         FIG. 14  is an alternate embodiment of the frequency response curve for the first filter k 1  shown in the block diagram of  FIG. 1 ; and 
         FIG. 15  is an alternate embodiment of the frequency response curve for the second filter k 2  shown in the block diagram of  FIG. 1 . 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     The present invention of a signal processing circuit, generally designated  40 , is shown in block diagram format in  FIG. 1 . An input signal is received at an input terminal  41  to a small magnitude output impedance stage  43 . Stage  43  preferably has an output impedance that is significantly smaller than the input impedance of a first filter k 1   44  so as not to materially affect the corner frequencies of the first filter  44 . First filter  44  is a complex filter with multiple user-adjustable corner frequencies and passband gains. The output of filter  44  is input into a first gain stage  45 . The output of the first gain stage  45  is input into a second gain stage  46 . The output of the second gain stage  46  is input into a second filter k 2   47 , which provides the output signal of the signal processing circuit  40  at a terminal  42 . 
     A preferred schematic for the signal processor circuit  40  is shown in  FIG. 2  with the blocks identified in  FIG. 1  shown in dashed lines about certain components of the schematic diagram. The design and operation of circuit  40  will now be further considered in its various portions corresponding to the blocks  43 - 47  shown in  FIGS. 1-2 . 
     In general an input signal, such as from a guitar, is buffered by the low output impedance stage  43  before presentation to the first filter  44 . For example, as shown in  FIG. 2 , the low output impedance stage  43  may consist of an amplifier  50  that is configured for unity gain. While not shown in block  43  of  FIG. 2 , it may also be desirable to provide low pass filtering at the input terminal  41 . For example, frequencies above the audio band, such as radio frequency interference (RFI) or the like, may be attenuated at or near the input to amplifier  50 . 
     First filter  44  shown in  FIG. 3  provides filtering of the low frequencies in the audio frequency range to prevent the generation of significant amounts of low frequency intermodulation (IMD) signals, which may result from the subsequent amplification by the first and second gain stages  45  and  46 . First filter  44  receives its input signal from the output of the low impedance stage  43  at an input terminal  51 . A resistor  52  and a capacitor  53 , connected in series, receive signals present on input terminal  51 . An opposite terminal of capacitor  53  is referenced to ground by a resistor  54 . 
     A single pole, multiple throw switch  55 , which may be a rotary switch with n positions, is connected to capacitor  53  and resistor  54 . Switch  55  selects one of n capacitors, such as capacitors  56 - 63  in the example shown in  FIG. 5 . Opposite ends of capacitors  56 - 63  are connected to a common node  65 . 
     A double pole, double throw switch  75  selects one of two networks that are also connected to node  65 . In the position shown in  FIG. 3 , switch  75  selects the first network that includes a pair of resistors  66  and  68 . Resistor  68  may be in the form of an adjustable resistor or potentiometer with an adjustable terminal  67  to control the amplitude of the signals provided through filter  44 . If switch  75  is in the opposite position from that shown in  FIG. 3 , the second network consisting of resistor  70 , capacitor  69  and variable resistor or potentiometer  73  is selected. This second network also provides control of the amplitude of the signals provided through filter  44  by varying the position of the adjustable terminal  72  of variable resistor  73 . In addition, capacitor  69  provides some additional filter effects over that of the first network consisting of resistors  66  and  68 . 
     Whichever network is selected by switch  75  provides the signals though the series connection of a capacitor  76  and a resistor  77  to the inverting input of an operational amplifier  80 . Op amp  80  has its non-inverting terminal referenced to ground. Op amp  80  also has a pair of diodes  81  and  82  oppositely poled between the output terminal and the inverting terminal of op amp  80  to keep op amp  80  from being overdriven. A resistor  84  and a capacitor  83  are also connected as feedback components, in parallel with diodes  81 - 82 , between the output terminal and inverting terminal of op amp  80 . Op amp  80  also provides the output signal of first filter  44  at an output terminal  85 . 
     First filter  44  provides different rates of signal gain or attenuation over different frequency ranges. In the illustrated embodiment of first filter  44 , there are four corner frequencies f 1 , f 2 , f 3  and f 4 , where each corner frequency is defined by the known equation f=1/(2πRC) and where R is the effective resistance at the frequency of interest, C is the effective capacitance at the frequency of interest and π is the well-known value of 3.1415 . . . . 
       FIGS. 4 through 7  illustrate the different effects that are provided by the first filter  44 . While  FIGS. 4-7 ,  10  and  12  do not have a scale along the frequency axis, it will be understood that these frequency response charts generally cover the frequency range of about 0 Hz to 20 KHz, which includes the audio frequency range, which is often specified as 20 Hz to 20 KHz. As will be presented more fully below, the frequency response of the first filter  44  depends upon which of capacitors  56 - 63  is selected by switch  55 , the first or second network selected by switch  75 , and the position or adjustment selected for potentiometers  68  or  73 . Irrespective of these selections, the gain versus frequency graphs shown in  FIGS. 4-7  will, in general, have a slope of 18 db/octave in a first frequency band from 0 Hz to f 1 , 12 db/octave in a second frequency band from f 1  to f 2 , 6 db/octave in a third frequency band from f 2  to f 3 , 0 db/octave in a fourth frequency band (which may also be referred to as a passband) from f 3  to f 4 , and −6 db/octave for frequencies above f 4 . 
     Filters, such as the first filter  44  that exhibits a slope of 18 db/octave in the lower frequency ranges and a passband of 0 db/octave in the higher frequency ranges are also known in the art as third order high pass filters. In the example of  FIG. 6 , there is additionally a high frequency rolloff of −6 db/octave above the corner frequency f 4 . Thus, a filter with the frequency response curve shown in  FIG. 4  could also be referred to as a third order high pass filter with high frequency rolloff. 
       FIG. 4  illustrates the effects of varying the passband gain with potentiometers  68  or  73 , depending upon which of the networks is selected by switch  75 . In frequency response graph  130 , the gain is set higher than in the graph  131 . Of course, if potentiometer  68  is set at for a higher gain value than potentiometer  73 , the user may switch from higher to lower gain (and, hence, from higher to lower volume) by changing switch  75  from the position shown in  FIG. 3  to the opposite position, and vice versa. To this end, switch  75  may be a foot-operated switch. As illustrated in  FIG. 4 , the changes in gain tend to have greater affect on those frequency bands that are less attenuated, such as those frequencies that lie between f 2  to beyond f 4 . If either of potentiometers  68 ,  73  are adjusted by moving the adjustable terminal  67  or  72  to its lower most position, the signal will be completely attenuated since lower pole of switch  75  is referenced to ground. Thus, potentiometers  68 ,  73  provide a broad range of signal attenuation. 
       FIG. 5  illustrates the ability to change the gain characteristics of those portions of the frequency response curve below frequency f 3 , including the frequency of the corner frequency f 3 . This is accomplished by changing the position of switch  55  to select one of capacitors  56 - 63 . Capacitors  56 - 63  are selected to be of different capacitive values to provide different frequency response characteristics.  FIG. 5  shows three different frequency response graphs  132 - 134  for three different capacitive values. Of course, with n capacitors of different capacitive value, n different frequency response curves will result instead of the three shown in  FIG. 5 . Note also that changing the capacitive value with switch  55  will also affect the corner frequency f 3 . In the example shown, corner frequency f 3   a  is associated with frequency response curve  132 , corner frequency f 3   b  is associated with frequency response curve  133  and corner frequency f 3   c  is associated with frequency response curve  134 . In general, a lower capacitive value for one of the capacitors  56 - 63  will cause the corner frequencies f 1 , f 2  and f 3  to shift toward higher frequencies. For example, in order to provide a range of effects through the selection of one of the n capacitors with switch  55  for audio signal applications, the capacitor with the lowest value preferably moves the 12 db/octave slope up to about 4 to 5 KHz. On the other hand, the capacitor with the highest capacitive value selected by switch  55  preferably moves the 12 db/octave slope down to about 30 Hz. Thus, the lower frequencies that the 12 db/octave portion of the frequency response curve operates on can range from about 30 Hz to about 5 KHz. The actual selection will depend upon the preferences of the user. 
       FIG. 6  illustrates the ability to change the gain characteristics of that portion of the frequency response curve above the corner frequency f 4 . The feedback components, capacitor  83  and resistor  84 , across op amp  80  normally determine the frequency of corner frequency f 4   a  when switch  75  is in the position shown in  FIG. 3 . This results in the frequency response graph shown by graph  136 . However, when switch  75  is in the opposite position to that shown in  FIG. 3 , capacitor  69  will change the frequency response to a graph such as graph  135  in  FIG. 6 . Note that in graph  135 , capacitor  69  also causes an increase in the corner frequency f 4   b  above that of f 4   a , and an increase in the higher frequency gain above that of graph  136 . 
       FIG. 7  is a composite of the frequency response graphs of  FIGS. 4-6 . The frequency shifts of some of the corner frequencies have not been illustrated, as in  FIGS. 4-6 , for purposes of simplifying this composite graph. It will thus be appreciated that the above-described differing techniques for customizing the frequency response characteristics of the first filter  44  provide the ability to customize or fine tune any portion of the audio frequency spectrum, as desired by the user. 
     The preferred embodiment of an amplifier for the first gain stage  45  in  FIG. 3  is shown in  FIG. 8 . An input terminal  88  of the first gain stage  45  passes input signals through a resistor  89  and a capacitor  90  to a node  97 . Node  97  is connected via a feedback resistor  91  to the output terminal of an op amp  98  and via a resistor  96  to the inverting input of op amp  98 . The non-inverting input of op amp  98  is referenced to ground. Feedback components, including a capacitor  94  and a resistor  95 , are connected from the inverting input to the output of op amp  98 . Oppositely poled diodes  92  and  93 , also connected from the inverting input to the output of op amp  98 , keep the op amp output amplitude limited. Diodes  92 - 93  clip symmetrically and therefore tend to limit the amount of distortion when the op amp  98  is overdriven. Diodes  92 - 93  also tend to provide some nonlinear distortion such as hysteresis when op amp  98  is overdriven since the feedback capacitor  94  will be charged by conduction of diodes  92 - 93 . However, when diodes  92 - 93  become non-conductive, the impedance seen by feedback capacitor  94  increases and capacitor  94  takes longer to discharge. Thus, the first feedback circuit consisting of diodes  92 - 93 , capacitor  94  and resistor  95  operates in two different impedance modes, depending upon whether diodes  92 - 93  are conductive or non-conductive. 
     The amplifier embodiment of  FIG. 8  has superior performance characteristics when used in signal processors for guitars. It is desirable for the best tonal characteristics resulting from clipping caused by gain stage  45 , when overdriven, that the clipping not be symmetrical. To this end, a second feedback circuit, consisting of resistors  89  and  91  and capacitor  90 , creates additional nonlinear distortion such as hysteresis in the response of the gain stage  45 . Resistor  96  provides some interaction between the first feedback circuit consisting of resistor  95 , capacitor  94  and diodes  92 - 93 , and the second feedback circuit. This additional nonlinear distortion such as hysteresis provides further distortion of the input signal by gain stage  45  when the op amp  98  is overdriven. 
     A simplified gain stage, generally designated  48 , is shown in  FIG. 9 , may be used in place of the gain stage  45  of  FIG. 8 , if desired. Simplified gain stage  48  is similar in structure and operation to gain stage  45 , except that resistors  91  and  96  of gain stage  45  that form a portion of an additional feedback loop about op amp  98  in  FIG. 8  are eliminated. Thus, the operation of gain stage  48  is similar in operation to the op amp  80  in the first filter  44 , as described above. 
     The gain stages employed in the second gain stage  46  in  FIG. 1  are preferably similar to those used in the first gain stage, and as shown in  FIG. 8  or  FIG. 9 . However, the second gain stage may have pairs of diodes  104 - 105  and  106 - 107  oppositely poled across the op amp  112  as shown in the complete schematic of  FIG. 2  to allow for greater amplitude signals before the diodes  104 - 107  become operative and limit the output amplitude. 
     Second gain stage  46  is connected in series or cascade with the first gain stage  45 . Each of gain stages  45 ,  46  preferably has a gain of greater than one and is nominally inverting. The frequency response for gain stages  45  or  46  is shown by a graph  137  in  FIG. 10 , and has a lower corner frequency f 1  and a higher corner frequency f 1 . From 0 Hz to f 1 , the slope is 6 db/octave. From f 1  to fh, which is the passband, the slope is 0 db/octave. At frequencies above fh, the slope is −6 db/octave. 
     The second filter stage, generally designated  47 , is shown in  FIG. 11 . An input terminal  116  receives input signals from the output terminal of the second gain stage  46 . Input terminal  116  is connected via a resistor  117  and capacitor  118  to a node  122 . A resistor  119  and a capacitor  120  are connected in series between node  122  and ground. Node  122  is also connected via a resistor  121  to another node  127 . A resistor  123  and a capacitor  124  are connected in series between node  127  and ground. Also separately connected in parallel between node  127  and ground are a capacitor  125  and a potentiometer  126 . The variable wiper arm of potentiometer  126  is connected to the output terminal  42  of the signal processor  40  of  FIG. 2 . Potentiometer  126  may function as the volume control for the signal processor. 
     The second filter  47  may have a complex frequency response as shown by the graph  138  in  FIG. 12 . Graph  138  may have six positive corner frequencies, f 5 , f 6 , f 7 , f 8 , f 9  and f 10 , in order of increasing frequency. From 0 Hz to corner frequency f 5 , the slope is 6 db/octave; from corner frequency f 5  to corner frequency f 6 , the slope is 0 db/octave; from corner frequency f 6  to corner frequency f 7 , the slope is −6 db/octave; from corner frequency f 7  to corner frequency f 8 , the slope is −12 db/octave; from corner frequency f 8  to corner frequency f 9 , the slope is −6 db/octave; from corner frequency f 9  to corner frequency f 10 , the slope is 0 db/octave; and above corner frequency f 10 , the slope is −6 db/octave. Capacitor  118  creates the low frequency rolloff below corner frequency f 5 , and capacitor  125  creates the high frequency rolloff above corner frequency f 10 . 
       FIG. 13  is a block diagram that is related to the block diagram shown in  FIG. 4 , but with the preferred frequency responses of the first and second filters  44 ,  47  shown in the filter blocks. In addition, the two gain stages  45 - 46  are shown combined in  FIG. 13  into a single stage. While preferred embodiments of the circuitry for the filters  44 ,  47  have been presented above in  FIGS. 3 and 11 , it will be appreciated by those skilled in the art that these filters could be active or passive and provide the desired frequency response curves. In accordance with one aspect of the present invention, at least 12 db/octave is used in the lower frequencies of the audio spectrum to provide greater attenuation of the lower audio frequencies. This helps minimize the production of lower frequency intermodulation distortion (IMD) frequency products, as previously discussed above, by the significant gain of the gain stages  45 - 46 . This avoids the commonly known muddy sound produced by prior art amplifiers. 
     The gain stages  45 - 46  may be combined into a single gain, or constitute a plurality of individual gain stages coupled together in the known cascade configuration. 
     The distortion produced may be modified by providing some offset voltage to the operational amplifiers, such as by referencing the non-inverting inputs to op amps  98  and  112  in FIGS.  2  and  8 - 9  to a reference (bias) voltage instead of to ground. Such use of bias voltage may be necessary if the op amps have unequal positive and negative supply voltages. These op amps  98  and  112  operate linearly so long as they are not overdriven. As previously discussed, if the op amps  98  and  112  are overdriven, the feedback diodes  92 - 93  and  104 - 107  will be rendered conductive. Thus, in the preferred embodiment of the invention, non-linearity of the gain stages results when these normally nonconductive diodes become conductive. These non-linearities may be modified, if desired, by offset biasing of the op amps  98  and  112 , such as by biasing the non-inverting inputs at a nonzero reference voltage. 
     An alternative frequency response curve  141  is shown in  FIG. 14  for the first filter  44 , instead of the frequency responses shown in  FIGS. 4-7 . In this embodiment, frequency response curve  141  has a slope of 12 db/octave at the lowest frequencies instead of 18 db/octave below the corner frequency f 1  in  FIGS. 4-7 . Curve  141  also does not have the high frequency rolloff of −6 db/octave for the higher frequencies, such as above the corner frequency f 4  in  FIGS. 4-7 . Characteristics of curve  141  can be provided by eliminating capacitors  53  and  83  in the schematic of filter  44  in  FIG. 3 . For example, short circuiting of capacitor  53  will eliminate the additional 6 db/octave of slope at the lowest frequencies of interest, thereby also eliminating the corner frequency f 1 . Elimination of capacitor  83  will also eliminate the corner frequency f 4  in  FIGS. 4-7  and the −6 db/octave rolloff for frequencies above f 4 . However, since capacitor  83  also contributes to the stability of op amp  80 , it may be desirable to simply decrease the capacitive value of capacitor  83  such that the corner frequency f 4  is above the frequencies of interest, and which effectively increases the passband of 0 db/octave slope. A first filter  44  with the frequency response characteristics of  FIG. 14 , instead of with the frequency response characteristics of  FIGS. 4-7 , will provide sufficient attenuation of the lower frequencies prior to amplification by the gain stages  45 - 46  to minimize IMD frequency products in many applications. 
     An alternative frequency response curve  142  is shown in  FIG. 15  for the second filter  47 , instead of the frequency response curve  138  shown in  FIG. 12 . In this embodiment, frequency response curve  142  has a slope of 0 db/octave at the lowest frequencies instead of 6 db/octave below the corner frequency f 5  in  FIG. 12 . Curve  142  also does not have the high frequency rolloff of −6 db/octave for the higher frequencies, such as above the corner frequency f 10  in  FIG. 12 . A filter having the frequency response curve shown in  FIG. 15  is known as a low pass filter. If the slope above the low frequencies is −12 db/octave for n=2, the filter may be referred to as a second order low pass filter. 
     The frequency response curve  138  in  FIG. 12  may be easily modified to resemble the frequency response curve  142  in  FIG. 15  by eliminating the low frequency rolloff capacitor  118  from the schematic shown in  FIG. 11  and by eliminating the high frequency rolloff capacitor  125 . This will also eliminate the corner frequencies f 5  and f 10  shown in  FIG. 12 . Alternately, capacitor  125  may be decreased in value such that the corner frequency f 10  is moved to a higher frequency beyond the frequency range shown in  FIG. 12 . 
     While particular embodiments of the invention have been shown and described, it will be obvious to those skilled in the art that changes and modifications may be made therein without departing from the invention in its broader aspects.

Technology Classification (CPC): 7