Patent Abstract:
A high-power modulation system includes drive circuitry that receives input signals from the signal source via a series of transformers. The drive circuitry amplifies the input signals and provides the resulting amplified signals to the high-power switch. The switch includes a series of stacked switching elements, each with a control terminal, first and second current-handling terminals, and feedback path extending between the first current-handling terminal and the control terminal. The feedback paths work in concert to turn the switches on and off together to prevent excessive voltage from developing across one or a subset of the switching elements. The feedback path includes a resistor that dampens the bandwidth of the feedback path to reduce turn-off and turn-on ringing and oscillation. The damping resistor may be coupled in series with a diode that holds charge against the control terminal of the switching element.

Full Description:
FIELD OF THE INVENTION 
     This invention relates to high-power, high-voltage modulators. 
     BACKGROUND 
     A broad range of applications require modulators and variable-voltage sources with high peak-power capabilities. Such applications include radar transmitters, X-ray machines, microwave-tube test sets, and semiconductor wafer manufacturing equipment. These machines and equipment employ such high-power amplifiers as cross-field amplifiers, traveling-wave tubes, magnetrons, klystron amplifiers (collectively referred to as “vacuum-electron devices”), and ion implanters. A number of high-voltage modulators are adapted to deliver pulsed power to these types of high-power amplifiers. 
     Conventional high-voltage modulators can be implemented using vacuum tubes, but the technology increasingly employs solid-state switches, which have higher peak-power capabilities and are more readily available. High-voltage modulators that employ solid-state switches provide excellent high-power, high-speed switching performance. There is always room for improvement, however, as competitive technology markets are ever watchful for cost-competitive systems that offer improved efficiency, reliability, speed performance, or a combination of these. For a detailed discussion of a conventional high-power modulator that employs solid-state switches, see U.S. Pat. No. 6,246,598,which is incorporated herein by reference. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The subject matter disclosed is illustrated by way of example, and not by way of limitation, in the figures of the accompanying drawings and in which like reference numerals refer to similar elements and in which: 
         FIG. 1  depicts a high-voltage modulator  100  driving an anode of a klystron K 1  in accordance with one embodiment. 
         FIG. 2  details embodiments of driver  120  and switch SW 2  of  FIG. 1 . Driver  120  is an “H” bridge with two identical halves driving respective inputs OFF and ON to the primary winding of transformer TX 2 . 
     
    
    
     DETAILED DESCRIPTION 
       FIG. 1  depicts a high-voltage modulator  100  driving an anode of a klystron K 1  in accordance with one embodiment. Modulator  100  drives klystron K 1  by alternately coupling the klystron anode to a high negative voltage via switch SW 1  and a low voltage via switch SW 2 . Each of switches SW 1  and SW 2  may include from one to N switch boards SB, coupled in series, with the number depending upon the magnitude of the switched voltage and the voltage rating of the switching elements on boards SB. A pair of driver boards DB 1  and DB 2  selectively turns switches SW 1  and SW 2  on and off responsive to optical trigger signals TRIG 1  and TRIG 2 . The driver boards are identical in this embodiment, as are the switches, so the following discussion is limited to driver board DB 1  and switch SW 1 . 
     Driver board DB 1  is powered by an AC source  105  via a first transformer TX 1  in this embodiment. The output terminals of transformer TX 1  connect to a conventional rectifier  110 , the positive and negative output terminals of which supply power to a controller block  115  and a driver  120  as a positive supply voltage VDR on a like-named supply node and zero volts on a ground node G 1 . Voltage VDR may vary, but is about fifteen volts in one embodiment. As with other designations used herein, VDR refers both to a signal and an associated node: whether a given reference is to a signal or a corresponding node will be clear from the context. 
     Driver  120  of driver board DB 1  provides output pulses to the primary windings of three transformers TX 2  via a pair of driver output terminals OFF and ON. The primary windings of transformers TX 2  are coupled in parallel so that the secondary windings issue simultaneous on and off pulses in response to pulses on output terminals OFF and ON. The pulses on the secondary windings of transformers TX 2  control switch boards SB to open and close switch SW 1 . In one embodiment, driver  120  issues a one microsecond pulse, from zero volts to VDR, on terminal ON to turn on switch SW 1 , and issues a subsequent similar pulse on terminal OFF to turn off switch SW 1 . The time between the pulses on node ON and node OFF determines the on and off times of switch SW 1 . 
     In one embodiment, switch SW 1  can be turned on for e.g. 100 us by asserting trigger signal TRIG 1  for 100 us: controller  115  issues a one-microsecond pulse on node PHB when signal TRIG 1  is asserted, and issues a subsequent one-microsecond pulse on node PHA when signal TRIG 1  is deasserted. Driver  120  responds to the pulses on nodes PHB and PHA by issuing corresponding pulses on respective driver output nodes ON and OFF to convey current through the primary winding of transformer TX 2 , and consequently through the secondary windings as well. The current through the secondary windings turns switch SW 1  on or off, depending upon the direction of current flow in the secondary windings. 
     The turns ratio of transformer TX 2  can vary, but is 1:1 in this example, with a single primary winding extending through the center of a toroid core. A single primary winding advantageously provides a high degree of voltage isolation between switch SW 1  and board DB 1 . The use of optical triggers and transformer TX 1  additionally isolates board DB 1 . 
     In some embodiments switch SW 1  may require one or more refresh pulses to remain on or off for a desired timing interval. In the embodiment of  FIG. 2 , discussed below, the switch may require refresh-on pulses to remain on for extended periods, but does not require refresh-off pulses. Refresh pulses can be initiated by trigger signals. Alternatively, controller  115  can be configured to refresh periodically to maintain a desired switch state. Other embodiments will differ, as will be readily understood by those of skill in the art. 
     Modulator  100  may be adapted to switch very high voltages. In one embodiment, klystron K 1  employs a cathode voltage VCAT of about −43 KV and a collector voltage of about zero volts. A cutoff supply  125  increases the absolute value of the anode voltage ANODE of klystron K 1  to 2 KV above cathode voltage VCAT (to −45 KV) for application by switch SW 1 . This 2 KV increase ensures klystron K 1  turns off and stays off when switch SW 1  closes. 
     Switching high voltages generates high charging currents through stray capacitances CS 1  and CS 2 , which can cause considerable electrical noise to couple to driver board DB 1 . (Other stray capacitances, such as those associated with the other transformers TX 2 , are omitted for ease of illustration.) Capacitors C 1  and C 2  are coupled between respective inputs of transformer TX 2  and a second ground G 2  to convey this switching noise to the primary winding of transformer TX 1  via stray capacitance CS 3  through transformer TX 1 . Ground G 2  may be inductively isolated from ground G 1  to prevent the noise coupled from switch SW 1  via transformer TX 2  from interfering with the operation of controller  115  and driver  120 . In one embodiment ground G 2  is isolated from driver board DB 1  by tying the ground side of capacitors C 1  and C 2  to a ground lug via a low-inductance conductor. 
       FIG. 2  details driver  120 , one of transformers TX 2 , and switch boards SB (all  FIG. 11 ) in accordance with one embodiment. In this example, driver  120  is an “H” bridge with two identical halves driving respective inputs OFF and ON to the primary winding of transformer TX 2 . Switch board SB includes a stack of e.g. twenty series-coupled switching elements SWE, each of which drops about 1/20 th  of the voltage across the one switch board SB. There being three switch boards in the embodiment of  FIG. 1 , each switching element SWE drops about 1/60 th  of the 45KV modulated on node ANODE in that embodiment. Each switching element SWE is disposed between and selectively couples first and second high-current source nodes SN 1  and SN 2 , and the depicted series of switching elements work in concert to selectively couple nodes IN and OUT. The configuration of each switching element is identical in this embodiment, so only the lowermost instance of switching element SWE is described in detail. The number of switching elements connected in series may change based upon the magnitude of the switched voltage and the voltage rating of the switching transistors. In one embodiment, the switching elements SWE employ insulated-gate bipolar transistors (IGBTs), but other types of power-switching devices might also be used, as will be readily understood by those of skill in the art. 
     The following discussion describes how driver  120  responds to a one-microsecond pulse (zero to VDR to zero) on node PHB by issuing a negative-going pulse (VDR to zero to VDR) on terminal ON. The pulse on terminal ON causes current to flow through the primary winding of transformer TX 2 . The resulting currents through the secondary windings of transformer TX 2  turn on all the switching elements SWE in the stack, and consequently turn on switch SW 2 . 
     To begin, assume switch SW 2  is off (not conducting) and both input signals PHA and PHB are at ground potential. In that case, signals PHA and PHB pull the control terminals of transistors Q 1  and Q 3  away from VDR, turning transistors Q 1  and Q 3  on and Q 2  and Q 4  off. Both input terminals to transformer TX 2  are therefore pulled to supply voltage VDR and no current flows through the primary winding. The breakdown voltages of zener diodes D 14  and D 16  are each 8.2 volts in an embodiment in which VDR is fifteen volts. Capacitors C 4  and C 3  bypass respective zeners D 14  and D 16  to reduce the time delay associated with turning on and off transistors Q 1  and Q 3 . 
     To initiate an on pulse, and thus close switch SW 2 , signal PHB is pulled to voltage VDR and signal PHA is left at ground potential. Raising PHB turns transistor Q 3  off and Q 4  on, creating a current path through transistor Q 1 , the primary winding of transformer TX 2 , and transistor Q 4 . To terminate the on pulse, signal PHB is returned to ground, which turns transistor Q 3  on and Q 4  off, eliminating the current path through the primary winding of transformer TX 2 . 
     A zener D 17  and a capacitor C 6  couple node PHB to the control terminal of transistor Q 4  in the same manner diode D 16  and capacitor C 3  couple node PHB to the control terminal of transistor Q 3 . Diodes D 8  and D 9  and resistors R 47  and R 48  ensure transistors Q 3  and Q 4  turn off faster than they turn on to prevent transistors Q 3  and Q 4  from conducting simultaneously during switching. A second pair of diodes D 11  and D 19  provide source-to-gate over-voltage protection for transistors Q 3  and Q 4 . The H bridge of driver  120  is unusual in that high-side transistors Q 1  and Q 3  are both biased on while awaiting an input signal on nodes PHA or PHB. The H bridge can therefore generate on pulses by turning on just one transistor. The same is true for off pulses, as noted below. Leaving on just the high-side transistors (or just the low-side transistors) may improve noise immunity by providing low-impedance paths from nodes ON and OFF and supply terminals that evince relatively stable voltages. A more traditional H bridge configuration may be used in other embodiments. 
     The current pulse from node OFF to node ON causes transformer TX 2  to send a current pulse through each secondary winding. With reference to the lowermost winding and the associated switching element SWE, the secondary current develops a positive voltage of e.g. fifteen volts across a resistor R 50 , the terminals of which are coupled to the input nodes to switch SW 2 . This voltage is transmitted to the gate of a high-voltage IGBT Q 5  via a resistor R 55  and a zener diode D 50 , thereby turning IGBT Q 5  on. Diode D 50  holds the resulting charge on the gate of IGBT Q 5  to keep the IGBT on after the voltage across resistor R 50  dissipates. The remaining series-coupled switching elements SWE likewise turn on, effectively closing switch SW 2 . 
     The charge collected on the gate of IGBT Q 5  bleeds off via a resistor R 60  and the leakage through a transient voltage suppressor T 1  that provides gate-to-source over-voltage protection for IGBT Q 5 . Unless this charge is refreshed, switch SW 2  will eventually shut off in this embodiment. It may therefore be necessary to refresh the charge periodically if the on-time of switch SW 2  is to be relatively long. The IGBTs will typically be turned on and refreshed such that the applied gate voltage maintains them in saturation for as long as switch SW 2  is closed. 
     To initiate an off pulse to open switch SW 2 , signal PHA to driver  120  is pulled to voltage VDR. Raising PHA turns transistor Q 1  off and Q 2  on, creating a current path through transistor Q 3 , the primary winding of transformer TX 2 , and transistor Q 2 . Returning signal PHA to ground potential turns transistor Q 1  on and Q 2  off, eliminating the current path through the primary winding of transformer TX 2 . The operation of the half of driver  120  disposed between nodes PHA and OFF is identical to the other half in the instant case, like-identified elements being the same or similar, so a detailed discussion is omitted for brevity. 
     The “off” current from node ON to node OFF causes transformer TX 2  to send a current pulse through each secondary winding in the opposite direction of on pulses. With reference to the lowermost switching element SWE, the current through the secondary winding develops a negative voltage of e.g. negative fifteen volts across resistor R 50 , which charges the gate of IGBT Q 5  in the opposite polarity through resistor R 55  and zener D 50 . In an embodiment in which zener D 50  drops ten volts, the gate-to-emitter voltages on each IGBT Q 5  in the series is about negative five volts. This gate-to-emitter voltage shuts off the transistors, effectively opening switch SW 2 . The off-pulses on the secondary windings of transformer TX 2  should be long enough to complete the transition of the associated transistors Q 5  from on to off. Likewise, the on-pulses on the secondary windings of transformer TX 2  should be long enough to complete the off-to-on transition. 
     A feedback path extending between the collector (first current-handling terminal) and gate (control terminal) of IGBT Q 5  includes a series of transient voltage suppressors T 2 , a diode D 55 , and a damping resistor R 65 . By elevating the gate voltage on transistor Q 5  when the collector voltage exceeds a predetermined level, the feedback path clamps the collector-to-emitter voltage of IGBT Q 5  to a level below the manufacturer&#39;s absolute maximum voltage rating. By adding or subtracting from the number of transient voltage suppressors T 2 , the clamping voltage between the collector and emitter of IGBT Q 5  can be adjusted to accommodate devices with different collector-to-emitter voltage ratings. 
     When the voltage between the collector and gate of IGBT Q 5  increases above the predetermined level, transient voltage suppressors T 2 , diode D 55 , resistor R 65 , and zener D 50  conduct current to the gate of IGBT Q 5  to keep the IGBT out of the cutoff mode. Keeping the IGBT out of the cutoff mode lowers the dynamic impedance of the IGBT, and hence the collector-emitter voltage across the IGBT. This action protects the IGBT from over-voltage conditions that might occur due to power-supply transients and when attempting to turn switch SW 2  on or off, particularly in the presence of an inductive load. Should transistor Q 5  in one switching element turn on more slowly than the others, the resulting voltage developed between the collector and the gate of the slower one of transistors Q 5  causes the feedback path to conduct charge to the gate of that transistor, and thus reduces the off-to-on transition time of the transistor. Similarly, should one of the series of transistors Q 5  turn off more slowly than the others, the resulting voltage developed between the collector and the gate of the transistor causes the feedback path to conduct charge to its gate to prevent the transistor from turning off too quickly. The feedback paths thus equalize the turn-off times and the turn-on times of switching elements SWE so that no switch or subset of switches suffers a potentially damaging over-voltage condition. This feature advantageously simplifies the task of matching the turn-on and turn-off times of the series devices. 
     Diode D 55  ensures that a majority of the on-current from the secondary winding of transformer TX 2  is delivered to the gate/emitter junction of the IGBT, rather than to charge the capacitance associated with the protection path, and additionally prevents current from passing from gate to collector when transistor Q 5  is in saturation. Resistor R 65  lowers the clamping response of collector-to-gate feedback path to reduce ringing and oscillation. The above-referenced U.S. Pat. No. 6,246,598 (the &#39;598 patent) describes a high-voltage modulator with a feedback scheme similar to that of  FIG. 2  of the instant application. The feedback path of the &#39;598 patent is coupled between the collector and gate of the associated transistor without the intervening diode D 50 , however. Diode D 50  isolates the gate of IGBT Q 5  from leakage current through the feedback path, as such leakage can interfere with the proper operation of the IGBT. Instead of developing a voltage across resistor R 60 , leakage current from the feedback path is dissipated through resistors R 55  and R 50 . The feedback path of the &#39;598 patent also lacks resistor R 65  and the associated damping effect. 
     Each IGBT has an associated series of balancing resistors R 70  between its emitter and collector. Balancing resistors R 70  ensure that each IGBT has about the same collector-to-emitter voltage when switch SW 2  is off. In some embodiments, a capacitor can be added in parallel with resistor R 70  to further dampen voltage transients, as shown in the &#39;598 patent. A diode D 60  connected in parallel with resistors R 70  is a fast-recovery voltage device that may be included to prevent transistor Q 5  from being reverse biased from emitter to collector. Such reverse conduction might occur, for example, if modulator  100  is used with an inductive load. 
     In the foregoing description and in the accompanying drawings, specific terminology and drawing symbols are set forth to provide a thorough understanding of the present invention. In some instances, the terminology and symbols may imply specific details that are not required to practice the invention. For example, signals described or depicted as having active-high or active-low logic levels may have opposite logic levels in alternative embodiments. As another example, circuits described or depicted as including IGBTs may alternatively be implemented using any other technology in which a signal-controlled current flow may be achieved. 
     While the present invention has been described in connection with specific embodiments, variations of these embodiments will be obvious to those of ordinary skill in the art. For example,
         1. while the circuitry employed in the feedback path of the above-described switching elements are implemented using discrete components, the switching and feedback circuitry may be integrated in other embodiments; and   2. switching elements and switch boards can be combined in parallel for increased current-handling. Moreover, some components are shown directly connected to one another while others are shown connected via intermediate components. In each instance the method of interconnection, or “coupling,” establishes some desired electrical communication between two or more circuit nodes, or terminals. Such coupling may often be accomplished using a number of circuit configurations, as will be understood by those of skill in the art. Therefore, the spirit and scope of the appended claims should not be limited to the foregoing description. Only those claims specifically reciting “means for” or “step for” should be construed in the manner required under the sixth paragraph of 35 U.S.C. Section 112.

Technology Classification (CPC): 7