Patent Abstract:
A system and method are provided for delivering power to a dynamic load. The system includes a power supply providing DC power having a substantially constant power open loop response, a power amplifier for converting the DC power to RF power, a sensor for measuring voltage, current and phase angle between voltage and current vectors associated with the RF power, an electrically controllable impedance matching system to modify the impedance of the power amplifier to at least a substantially matched impedance of a dynamic load, and a controller for controlling the electrically controllable impedance matching system. The system further includes a sensor calibration measuring module for determining power delivered by the power amplifier, an electronic matching system calibration module for determining power delivered to a dynamic load, and a power dissipation module for calculating power dissipated in the electrically controllable impedance matching system.

Full Description:
RELATED APPLICATION  
       [0001]     This application claims the benefit of U.S. Provisional Application No. 60/731,797, filed on Oct. 31, 2005, the entire teachings of which are incorporated herein by reference.  
       BACKGROUND  
       [0002]     Various approaches exist for providing RF power to dynamic loads. RF generators provide power to dynamic loads typically at frequencies between about 400 kHz and about 200 MHz. Frequencies used in some scientific, industrial and medical applications are approximately 2 MHz, 13.56 MHz and 27 MHz.  
         [0003]     As shown in  FIG. 1A , one system  100  for providing RF power to dynamic loads (i.e., a plasma load  140 ) involves a fixed frequency RF generator  110  and a two-axis tunable matching network  120  connected by a 50 Ω transmission line  130 . The tunable matching network  120  includes a series motorized vacuum variable capacitor  122  and inductor  124  and a shunt motorized vacuum variable capacitor  126 . The algorithm used to determine the series and shunt capacitance is based on impedance measurements typically made using a magnitude and phase detector  150 . Independent power control is based on power measurements at the RF generator  110 . The power control loop  160  and impedance control loop  162  are independent.  
         [0004]     As shown in  FIG. 1B , another system  100 ′ for providing RF power to dynamic loads involves a fixed element matching network  120 ′ fed by an RF generator  110  and connected by a 50 Ω transmission line  130 . The fixed element matching network  120 ′ includes a series capacitor  122  and inductor  124  and a shunt capacitor  126 . The frequency of the RF generator  110  can be tuned to a certain range (e.g., 13.56 MHz ±5%). The RF generator  110  frequency command is based on the value of voltage standing wave ratio (VSWR). The independent power loop and VSWR (impedance) control loop  160 ′ are based on measurements at the output of the RF generator  110 .  
         [0005]     As shown in  FIG. 1C , another system  100 ″ for providing RF power to dynamic loads involves an integrated RF generator-impedance matching network  120 ″. The RF generator-impedance matching network  120 ″ includes a series capacitor  122  and inductor  124  and a plurality of shunt capacitor  126   a  . . .  126   n . The shunt capacitor  126   a  . . .  126   n  are coupled to a switching circuit  127   a  . . .  127   n  that couples and decouples the capacitors  126  to ground. The power control and frequency control  160 ″ of the system  100 ″ are not conducted simultaneously.  
       SUMMARY  
       [0006]     These prior art techniques and methods have disadvantages. Higher cost is typically associated with prior art techniques and methods due to the need for at least two separate modules: 1) the RF generator/amplifier and 2) the impedance matching network, which are to be connected via a transmission line. Furthermore, each module requires a RF voltage/current sensor or a magnitude/phase detector.  
         [0007]     Plasma impedance is a function of the power delivered to the plasma. Furthermore, the power delivered by the RF generator is a function of the impedance “seen” by the generator. As a result, a clear circular interdependence exists between delivered power and load impedance yielding a multi-input-multi-output (MIMO) system with cross-coupling. In prior art systems, the RF generator control loop and the impedance matching control loop are independent and thus cannot compensate for the cross-coupling between power control and impedance matching control loops. This leads to poor closed-loop performance.  
         [0008]     The dynamic response of any controlled system is only as fast as the slowest functional module (sensor, actuator, or control system parameters). In prior art systems, the slowest functional module is typically the DC power supply. Specifically, the DC power supplied to the input of the RF power amplifier usually includes a large electrolytic capacitor that is used to filter higher frequencies. The downside of using such a filter network is that the dynamic response (e.g., response to a step change in power command) is slow regardless of the control update rate. The system is therefore unable to sufficiently compensate for plasma instabilities.  
         [0009]     In systems that use a vacuum capacitor driven by motors, the response time is on the order of hundreds of milliseconds. Owing to the fact that plasma transients (sudden and rapid change of impedance) of interest occur within hundreds of microseconds, the vacuum capacitor cannot be used to match load changes attributed to plasma transients.  
         [0010]     Control algorithms for matching networks used in the prior art have relied upon the real and imaginary components of the measured impedance. Impedance measurement-based matching control suffers from an inherent disadvantage. For example, a change in shunt capacitance to correct or modify the real component of the impedance results in an undesirable change in the imaginary component of the impedance. Similarly, a change in the series capacitance or frequency to correct or modify the imaginary component of the impedance results in an undesirable change in the real component of the impedance. The matrix that relates the controlled variable vector (formulated by the real and imaginary components of the impedance) and the controlling variable vector (formulated by the shunt and series capacitance or the shunt capacitance and frequency) is non-diagonal. Impedance measurement-based control algorithms are therefore not effective. Control algorithms based on the impedance formulated by using magnitude and phase measurements of the impedance are similarly ineffective.  
         [0011]     Calibration methods for prior art systems calibrate the RF impedance analyzer or VI probe at the input of the electronic matching network. These calibration methods assume the power loss in the electronic matching network is fixed for all states of the electronic matching network and operating frequencies. However, the losses of the electronic matching network contribute significantly to the overall system operation.  
         [0012]     Accordingly, a need therefore exists for improved methods and systems for controlling power supplied to a dynamic plasma load and the losses associated therewith.  
         [0013]     There is provided a system for delivering power to a dynamic load. The system includes a power supply providing DC power having a substantially constant power open loop response, a power amplifier for converting the DC power to RF power, a sensor for measuring voltage, current and phase angle between voltage and current vectors associated with the RF power, an electrically controllable impedance matching system to modify the impedance of the power amplifier to at lease substantially match an impedance of a dynamic load, and a controller for controlling the electrically controllable impedance matching system. The system further includes a sensor calibration measuring module for determining power delivered by the power amplifier, an electronic matching system calibration module for determining power delivered to a dynamic load, and a power dissipation module for calculating power dissipated in the electrically controllable impedance matching system.  
         [0014]     In one embodiment, the electrically controllable impedance matching system can include an inductor, a capacitor in series with the inductor, and a plurality of switched capacitors in parallel with the dynamic load. The inductor can be a multiple tap-type inductor or a variable-type inductor. Each of the plurality of switched capacitors can be in series with a switch and an additional capacitor. In another embodiment, the electrically controllable impedance matching system can include a capacitor, and a plurality of switched capacitors in parallel with the dynamic load, wherein each of the plurality of capacitors is in series with a switch and an additional capacitor. In yet another embodiment, the electrically controllable impedance matching system can control the frequency of the impedance matching between the power amplifier and the dynamic load.  
         [0015]     In one embodiment, the controller can control the electrically controllable impedance matching system for simultaneous control of conductance and susceptance associated with the impedance between the power amplifier and the dynamic load. In another embodiment, the controller can simultaneously control RF power frequency, RF power magnitude and the impedance between the power amplifier and the dynamic load. In yet another embodiment, the controller can control the electrically controllable impedance matching system for regulating conductance and susceptance to setpoints that stabilize an unstable dynamic load.  
         [0016]     The power dissipated in the electrically controllable impedance matching system is the difference between the power delivered by the power amplifier and the power delivered to the dynamic load. The power delivered to the dynamic load is a sum of the power delivered to a resistive load and the power dissipated inside the load simulator.  
         [0017]     The sensor calibration measuring module calibrates the sensor into a resistive load, wherein the resistive load is 50 Ω. The electronic matching module calibrates an output of the electrically controllable impedance matching system into a load simulator. The load simulator can be an inverse electrically controllable impedance matching system. The electronic matching system calibration module can include a power meter calibration module for determining power delivered to a resistive load; and a load simulator calibration module for determining power dissipated inside the load simulator. The resistive load can be 50 Ω. The radio frequency power delivery system provides at least the following advantages over prior art systems. The system can enhance power setpoint regulation, impedance matching, and load disturbance mitigation using high-speed (e.g., in excess of 50 kHz in one embodiment) digital multi-input-multi-output (MIMO) control. The system can operate in the presence of transient changes in plasma load properties and under conditions involving fast plasma stabilization. The system can provide a RF power delivery system that is robust to transients during startup of the system. The system can provide a high power step-up ratio, wherein the high power step-up ratio is 100 (e.g., 15 W to 1500 W). The system can measure power delivered to the load connected to the output of the integrated generator system. The system can allow for regulation of power that is independent of the power loss variation associated with the state/value of various controlled variables. The system can eliminate the need for recipe-based calibration for plasma loads. 
     
    
     BRIEF DESCRIPTIONS OF THE DRAWINGS  
       [0018]     The foregoing and other objects, features and advantages of the invention will be apparent from the following more particular description of preferred embodiments of the invention, as illustrated in the accompanying drawings. The drawings are not necessarily to scale, emphasis instead being placed upon illustrating the principles of the invention.  
         [0019]      FIG. 1A  is a diagram of an RF power delivery system having a two-axis tunable matching network according to the prior art;  
         [0020]      FIG. 1B  is a diagram of an RF power delivery system having a fixed matching network according to the prior art;  
         [0021]      FIG. 1C  is a diagram of an RF power delivery system having an integrated RF generator-impedance matching network according to the prior art;  
         [0022]      FIG. 2  is a module-based diagram of the On-Chamber RF power delivery system;  
         [0023]      FIG. 3  is a plasma stability graph;  
         [0024]      FIG. 4  is one embodiment of a fast DC bus of  FIG. 2 ;  
         [0025]      FIG. 5  is one embodiment of an RF impedance analyzer or VI Probe of  FIG. 2   
         [0026]      FIG. 6  is one embodiment of an electronic matching network of  FIG. 2 ;  
         [0027]      FIG. 7  is one embodiment of a module-based diagram of a DSP compensator board of  FIG. 2 ;  
         [0028]      FIG. 8  is a block diagram for calibrating the On-Chamber RF power delivery system;  
         [0029]      FIG. 9A  is one embodiment for calibrating a power meter to a 50 Ω calorimeter power reference;  
         [0030]      FIG. 9B  is one embodiment for calibrating a load simulator to a DC power reference;  
         [0031]      FIG. 9C  is one embodiment for calibrating an RF impedance analyzer into a 50 Ω load; and  
         [0032]      FIG. 9D  is one embodiment for calibrating power delivered into the load simulator. 
     
    
     DETAILED DESCRIPTION  
       [0033]     Generally, an integrated radio frequency (RF) power delivery system is provided for dynamic load applications (e.g., inductive and/or capacitive plasma load).  FIG. 2  is an illustration of the integrated radio frequency (RF) power delivery system  200 . Representative functional modules of the integrated system  200  include a fast DC bus  210 , an RF power amplifier (“PA”)  220 , a digital signal processor (“DSP”) compensator board  230 , an RF impedance analyzer or VI probe  240 , and an electronic matching network  250 . The system  200  is coupled to a plasma load  260 . It should be understood by one skilled in the art that the integrated system  200  can be implemented for a wide range of resistive and reactive loads.  
         [0034]     Generally, the fast DC bus  210  delivers DC power to the power amplifier  220 . The power amplifier  220  converts the DC power from the fast DC bus  210  to an RF power at a frequency. The electronic matching system  250  switches shunt capacitors (not shown) to match the impedance between the power amplifier  220  and the plasma load  260  to facilitate stable and maximum power transfer from the power amplifier  220  to the plasma load  260 . The DSP compensator board  230  controls the operation of the system  200  based on measurements received from the fast bus controller  212  and RF impedance analyzer  240 . The RF impedance analyzer  240  measures the RMS voltage, RMS current, and phase angle between the RF voltage and current vectors. Based on these measurements, relevant RF parameters are computed by the DSP compensator board  230 . These parameters include, but are not limited to impedance vector  z , admittance vector  y , delivered power P del , and voltage-standing wave ratio (“VSWR”). Typical operations of the DSP compensator board include power setpoints through the fast bus controller  212 , RF power frequency setpoints through the power amplifier driver  222 , and switching frequency through the electronic match controller  252 .  
         [0035]     In one aspect, the system  200  achieves simultaneous power and impedance regulation. Independent susceptance regulation allows for the implementation of a frequency control algorithm based only on the deviation of the conductance from the conductance setpoint. As a result, both control loops can be operated simultaneously and at high-speed resulting in improved robustness. Further, well-known instabilities for electronegative plasmas at low-pressure (e.g., SF 6  at 5 mT at 300 W as illustrated in  FIG. 3 ) can be stabilized by setting arbitrary conductance and susceptance setpoints in conjunction with operation of the Fast DC bus  210 .  
         [0036]      FIG. 4  is a diagram of a partial resonant inverter power supply type fast DC bus  210 . The fast DC bus  210  provides process stability due to its associated constant power open loop response. The fast DC bus  210  improves FET utilization over the entire load space which results in more power being delivered to the load with the same PA  220  ( FIG. 2 ). The fast DC bus  210  has a fast response rate allowing it to deliver increased power to the plasma so it does not extinguish while also allowing the flexibility to reduce the bus voltage to ensure the FETs on the PA  220  operate in a safe mode. Other types of topologies can for the fast DC bus  210  can be used. See for example, co-pending continuation-in-part application it&#39;s parent U.S. application Ser. No. 10/947,397 filed Sep. 22, 2004, the entire teaching of each application are herein incorporated by reference.  
         [0037]     In one embodiment, the fast DC bus can be a partial resonant inverter  210  that includes a pair of switches (MOSFETs)  302   a ,  302   b , an inductor (L)  306 , a capacitor (C)  308 , and four diodes  310   a ,  310   b ,  310   c , and  310   d . In operation, the partial resonant inverter  300  converts the input voltage into a square wave or other known type DC wave form. The square wave is passed through the inductor  306  and capacitor  308 , the combination of which form an LC filter, clamped by the diodes  310   c ,  310   d , coupled and rectified by a transformer rectifier  304  and filtered to obtain a desired DC voltage (power setpoint). The DC power setpoint is provided from the DSP compensator board  230  ( FIG. 2 ). The desired impedance setpoint can be specified in terms of its vector inverse (referred to as admittance) and which constitutes simultaneous regulation of conductance to an arbitrary conductance setpoint and regulation of susceptance to an arbitrary susceptance setpoint. The output of the partial resonant inverter  300  (DC-DC converter) is connected to DC input of the RF power generator/amplifier  220 .  
         [0038]     In operation, the capacitor  308  is periodically charged to an input rail voltage (+Vin) and discharged while the capacitor current is passed via the plasma load  260  ( FIG. 2 ). Every charge or discharge cycle, the energy deposited in the resistive load is equal to CV 2 /2, independent of load resistance. Thus, the power is equal to F SW ×CV 2 /2, where F SW  is the switching frequency and V is the input voltage. The inductor  306  ensures that the capacitor  308  is fully charged and discharged in finite time. One advantage of the partial resonant inverter  300  design is the ability to control the output voltage by varying either V or/and F SW .  
         [0039]      FIG. 5  is a diagram of one embodiment of an RF impedance analyzer or VI Probe  240 . The VI Probe  240  includes a DC power supply  242 , an analysis board assembly  244 , and a probe head assembly  246 . The analysis board assembly  244  receives low-level RF signals from the probe head assembly  246 . The probe head assembly  246  provides two voltage outputs: 1) a voltage representation of the time varying electric field present in the probe head assembly  246  (voltage signal); and 2) a voltage representation of the time varying magnetic field present in the probe head assembly  246  (current signal). The analysis board assembly  244  receives and processes the two voltage outputs of the probe head assembly  246  and outputs the RF parameters to the DSP compensator board  230  ( FIG. 2 ). MKS Instruments, Inc. VI-Probe- 4100  and VI-Probe- 350  are exemplary analyzers that can be used for this purpose.  
         [0040]      FIG. 6  is a diagram of one embodiment of an electronic matching network  250 . In one embodiment, the electronic matching  250  includes an inductance  254  in series with the load  260  (e.g., a compact inductor with multiple tap points), a fixed or variable series-padding capacitor  252 , and field effect transistors (“FET&#39;s”)  256   a  . . .  256   n  that switch one or more upper capacitors C tu (i)  258   a  . . .  258   n  to a corresponding lower capacitor C td (i)  258   a ′ . . .  258   n ′, which is terminated to ground. In some embodiments, the electronic matching  250  network does not include the inductance  254  in series with the load  260 . Other types of electronic matching networks can be used. See for example, U.S. Pat. No. 6,887,339, the entire teaching of which is herein incorporated by reference.  
         [0041]      FIG. 7  shows a module-based diagram of a DSP compensator board  230 . The DSP compensator board  230  incorporates both a digital signal processor (“DSP”) and a field programmable gate array (“FPGA”), and together controls the entire integrated system  200 . The DSP compensator board includes an admittance compensation module  232 , a frequency control module  234 , an electronic match control module  236 , an RF power computation module  237 , and an RF power control module  238 . Generally, the DSP compensator board receives the output from the VP probe  240 . The admittance computation module  232  uses the VI probe outputs to calculate the admittance of the system  200 . The frequency control module  234  uses the admittance to vary the frequency of the power amplifier  220 . The electronic match control module  236  uses the admittance to switch the FETs  256  of the electronic matching network  250  on or off. The RF power computation module  237  uses the VI probe outputs to calculate the RF power of the system  200 . The RF power control module  234  uses the RF power computation to regulate the power supplied from the fast DC bus power  210 . A more detailed description of the operation of the system  200  is set forth below.  
         [0042]     One embodiment of the power regulation objective and algorithm is set forth below: The objective is to regulate the delivered power P del  to a user-defined setpoint P sp . To ensure smooth transitions, trajectory generators are used. In one embodiment, a first-order trajectory is generated as follows:  
                 ⅆ     P   t         ⅆ   t       =       1     τ   t       ⁢     (         P   t     ⁡     (   t   )       -     P   sp       )               EQN   .           ⁢   1             
 
 where τ t  is the trajectory time constant and P t  is the desired power trajectory. The delivered-power control algorithm, in terms of the change in power commanded to the Fast Bus, is given by the following relationship:
 
 P   cmd   =k   p ( P   t   −P   del )+ k   i ∫( P   t   −P   del ) dt   EQN. 2
 
 where k P  and k i  are the proportional and integral gains, respectively. 
 
         [0043]     Admittance regulation objective: A normalized admittance vector is defined as follows:  y =g+jb where g is the normalized conductance and b is the normalized susceptance. The impedance matching control objective is formulated as follows: g→g sp  and b→b sp  where g sp  and b sp  are arbitrary setpoints selected to improve plasma stability. The above objective is reinterpreted in terms of impedance by noting that impedance is defined as the reciprocal of admittance, according to the following relationship:  
             z   =       1   y     =       r   +     j   ⁢           ⁢   x       =         R   +     j   ⁢           ⁢   X         Z   0       =       R   +     j   ⁢           ⁢   X           R   0     +     j   ⁢           ⁢   0                       EQN   .           ⁢   3             
 
 where z is the normalized impedance, r and x are the resistance and reactance, respectively, Z 0 =R 0 +j 0  denotes a nominal RF amplifier characteristic impedance. It follows that when g→1 and b→0, we obtain R→R 0  and X→0. 
 
         [0044]     Admittance regulation algorithm: The frequency control loop is designed by using conductance measurements, for example, as a PI control algorithm as follows:
 
 f   tcmd   =−k   pf ( g   sp   −g )− k   if ∫( g   sp   −g ) dt   EQN. 4
 
 where k pf  and k if  are scalar proportional and integral control gains. The shunt capacitance control loop is designed by using conductance measurements, for example, as a PI control algorithm as follows:
 
 C   tcmd   =−k   pc ( b   sp   −b )− k   ic ∫( b   sp   −b ) dt   EQN. 5
 
 where d pc  and k ic  are scalar proportional and integral control gains. 
 
         [0045]     In operation, referring now to  FIGS. 2, 3  and  6 , after the user provides a non-zero setpoint, the trajectory generator and the power and admittance control algorithms are simultaneously activated and executed. The VI probe  240  provides analog signals proportional to the RF voltage and RF current, which are synchronously sampled by the analog-to-digital converters, sent to a mixer and CIC filter (not shown) and ultimately sent through a calibration matrix to yield RF voltage and RF current measurements given by the following relationships:
 
    V   =V r   +jV   i 
 
 and 
 
    I   =I r   +jI   i   EQN. 6
 
 where  V ,  I  denote vector representations of the instantaneous RF voltage and current, respectively, and subscripts r and i are used to denote the scalar values of the real and imaginary components. 
 
         [0046]     The average delivered power is computed as follows:  
               P   del     =         1   2     ⁢   Re   ⁢     {       VI   _     *     }       =         V   r     ⁢     I   r       +       V   i     ⁢     I   i                   EQN   .           ⁢   7             
 
 where Re{} denotes the real component of the vector, and superscript * is used to denote the complex conjugate of the vector. 
 
         [0047]     The admittance vector  Y  is then computed as follows:  
               Y   _     =         I   _       V   _       =           (         I   r     ⁢     V   r       +       I   i     ⁢     V   i         )         V   r   2     +     V   i   2         +     j   ⁢       (         I   i     ⁢     V   r       -       I   r     ⁢     V   i         )         V   r   2     +     V   i   2             ≡     G   +     j   ⁢           ⁢   B                   EQN   .           ⁢   8             
 
 where the conductance G and the susceptance B are real and imaginary components of the admittance  Y . 
 
         [0048]     The normalized conductance g and nonnalized susceptance b are computed as follows:  
               g   =         Z   0     ⁢   G     =       Z   0     ⁢       (         I   r     ⁢     V   r       +       I   i     ⁢     V   i         )         V   r   2     +     V   i   2               ⁢           ⁢     
     ⁢   and   ⁢           ⁢     
     ⁢     b   =         Z   0     ⁢   B     =       Z   0     ⁢       (         I   i     ⁢     V   r       -       I   r     ⁢     V   i         )         V   r   2     +     V   i   2                       EQN   .           ⁢   9             
 
 where Z 0  denotes the characteristic impedance of the RF amplifier. The measurements of P del , g, b are respectively sent to the control algorithms for P cmd , f cmd , C tcmd  respectively. 
 
         [0049]     The electronic match controller  252  switches the FETs  256  ( FIG. 6 ) thereby switching the shunt capacitors  258  to match the impedance between the power amplifier  20  and the dynamic load  260 . The absence of moving mechanical parts leads to higher reliability. In one embodiment, the step response of the system  200  is faster than about 1 ms because the speed of the response is governed by the electronics and not by the mechanical response.  
         [0050]     A change in frequency results in a change in both the conductance and the susceptance. However, for an integrated system without transmission line cables, a change in shunt capacitance results only in a change in the susceptance and does not affect the conductance value. Thus, the matrix that relates the controlled variable vector (formulated by the real and imaginary components of the admittance) and the controlling variable vector (formulated by the shunt and series capacitance or the shunt and frequency) is triangular. As a result, independent susceptance regulation is achieved by varying the shunt capacitance.  
         [0051]     Independent susceptance regulation allows for the implementation of a frequency control algorithm based only on the deviation of the conductance from the conductance setpoint. As a result, both the conductance-based frequency control loop and the susceptance-based shunt capacitance control loop can be operated simultaneously and at high-speed, resulting in improved robustness.  
         [0052]      FIG. 8  is a block diagram  300  of a method for determining the power dissipated (loss) in the electronic matching network  250  ( FIG. 2 ) to improve the efficiency of the system  200 . Step one ( 310 ), a power meter  314  ( FIG. 9A ) is calibrated into a 50 Ω calorimeter power reference to determine the power delivered to the 50 Ω load. Step two ( 320 ), a load simulator calorimeter  332  ( FIG. 9B ) is calibrated to a DC power reference to determine the power dissipated inside a load simulator  342  ( FIG. 9D ). Step three ( 330 ), the VI probe  240  ( FIG. 2 ) is calibrated into a 50 Ω load to determine the power delivered by the power amplifier  220  ( FIG. 2 ). Step four ( 340 ), the output of the system  200  is calibrated into the load simulator  342  to determine the power delivered to Z L =R L +jX L . Step  5  ( 350 ), the power dissipated in the electronic matching system is calculated by difference between the he power delivered by the power amplifier  220  and the power delivered to. Z L =R L +jX L .  
         [0053]      FIG. 9A  is detailed implementation diagram of step  310  for calibrating the power meter  314 . A calorimeter  322  is coupled to the output of the VI Probe  240 , RF power is applied from the power amplifier  220 , and the power meter  314  is calibrated. Calorimetry is the measurement of thermal losses. It is implemented by thermally insulating the 50 Ω load in the calorimeter ( 322 ) to prevent ambient thermal losses and measuring the flow rate and the temperature rise of the cooling water. The power meter is calibrated to the power dissipation in the load computed by  
         Q   =         ⅆ   m       ⅆ   t       ⁢     C   ⁡     (       T   out     -     T     i   ⁢           ⁢   n         )           ,           ⁢     where   ⁢           ⁢       ⅆ   m       ⅆ   t               
 denotes the mass flow rate, C denotes the specific heat of water, and T in , T out  denote the inlet and outlet temperatures, respectively. A computer  324  acquires flow rate and temperature measurements to compute the power dissipation in the load and the difference (error) with respect to readout of the power meter. The computer  324  then applies this error as a correction to the power meter to complete the calibration. 
 
         [0054]      FIG. 9B  is detailed implementation diagram of step  320  for calibrating the load simulator calorimeter  332 . A load simulator calorimeter  332  is coupled to a DC power supply  334 , DC power is applied, and the load simulator calorimeter  332  is calibrated. The DC power supply provides the DC power measurements. Using flow rate and temperature measurements at the inlet and outlet of the cooling system, a computer  324  computes the power dissipated in the load simulator. The computer  324  then applies the error between the power reported by the DC power supply and the power computed using calorimetry as a correction to the load simulator to complete the calibration.  
         [0055]      FIG. 9C  is detailed implementation diagram step  330  for calibrating an RF impedance analyzer or VI probe  240 . Generally, the VI Probe  240  calibration in each integrated RF generator system  200  includes the following steps that yield a matrix transfer function that relates the VI probe voltage and current measured by the DSP compensator board  230  to an actual RF line voltage and current.  
         [0056]     First, a short circuit connector  312  is coupled to the RF line output terminal of the VI probe  240 , RF power is applied from the power amplifier  220 , and Z dsp   sc  is computed, wherein Z dsp   sc  is defined as the ratio of V dsp /I dsp  as measured by the DSP compensator board  230  for short circuit. Second, an open circuit connector  314  is coupled to the RF line output terminal of the VI probe  240 , RF power is applied from the power amplifier  220 , and Z dsp   ac  is computed, wherein Z dsp   ac  is defined as the ratio of V dsp /I dsp  as measured by the DSP compensator board  230  for open circuit. Third, a 50 Ω load (Z L )  316  is coupled to the output of the VI Probe  240 , RF power is applied from the power amplifier  220 , V m  and I m  are recorded and the RF line voltage V L  is computed, wherein V L =√{square root over (P L Z L )}·P L  is the delivered power measured by a power meter  318  at the 50 Ω load  316 . Lastly, the VI probe calibration matrix transfer function is computed by the following equation:  
               [         V   L     ⁡     (   t   )           I   L     ⁡     (   t   )         ]     =       (             V   L         V   m     -       Z   sc   dsp     ⁢     I   m                     -     Z   sc   dsp       ⁢     V   L           V   m     -       Z   sc   dsp     ⁢     I   m                       -     V   L           Z   L     ⁡     (         -     Z   sc   dsp       ⁢     I   m       -     V   m       )                   Z   sc   dsp     ⁢     V   L           Z   L     ⁡     (         -     Z   sc   dsp       ⁢     I   m       -     V   m       )               )     ⁡     [         V   dsp     ⁡     (   t   )           I   dsp     ⁡     (   t   )         ]               EQN   .           ⁢   10             
 
         [0057]     The expression in equation  10  translates VI probe measurement signals into RF line voltage and RF line current at the output of the VI probe  240 .  
         [0058]      FIG. 9D  is detailed implementation diagram step  340  for calibrating the system  200  ( FIG. 2 ). The system level calibration is used to quantify the power loss in the electronic matching network  250  for a range of values matching network variables. A load simulator  342  is coupled to the output of the electronic matching network  250 . Typically, the load simulator is an electronic matching network inverse to the electronic matching network  250 . A 50 Ω load is coupled to the output of the load simulator  342 . The system-level calibration of the RF generator system  200  is performed as follows. First, a series inductance is adjusted in ll steps for L S ε[L s min , L s max ]. Second, a power setpoint value is changed in pp steps P sp ε[P sp min , P sp max ] W. Third, a shunt capacitance setpoint value is changed in cc steps C tcmd ε[C tcmd min , C tcmd max ]. Lastly, an RF frequency value is changed in ff steps fε[f min , f max ] Hz.  
         [0059]     For each combination of the aforementioned steps, the load simulator  342  is set to present an impedance mismatch at the output of the electronic matching network  250 . Next, RF power is applied from the power amplifier  220  and the power meter  314  measures the terminating load  312  resistance. The terminating load resistance is denoted by P 50Ω  and transformed to the input of the load simulator  342 . The simulated load is denoted by P sys  as P sys =f 50-to-sim (P 50Ω , C 1 , C 2 ), where C 1 , and C 2 , represent the series and shunt capacitance of the load simulator and f 50-to-sim  represents a tabular arrangement. The losses associated in electronic matching network  250  is computed by the difference between the P L  and P 50Ω .  
         [0060]     In some embodiments, a calibration table which has dimensions ll×pp×cc×ff can stored in non-volatile memory (e.g., flash memory) as P sys =f VI-to-sim (L s , P sp , C tcmd , f), where f VI-to-sin  represents a tabular arrangement. High-speed real-time control loops necessitate fast searches through the calibration table during operation of the system  200 . Non-volatile memory (e.g., flash memory) tends to be slower than the volatile memory (e.g., Dynamic RAM). The high-speed volatile memory is effectively utilized, wherein the arrangement of the calibration table (dimensions ll×pp×cc×ff) can be based on how frequently L s , P sp , C tmcd , and f are changed. Specifically, the calibration table can be segmented into ll memory blocks; each block including pp memory pages; each memory page including a cc×ff dimensional table. A new memory block can be loaded into non-volatile memory when L s  is changed, a new memory page can be loaded when power setpoint is changed, and calibration points for the appropriate memory page associated with C tcmd  and f can be executed in real-time.  
         [0061]     While this invention has been particularly shown and described with references to preferred embodiments thereof, it will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the scope of the invention encompassed by the appended claims.

Technology Classification (CPC): 7