Patent Abstract:
A power converter includes a power transformer, primary-side circuitry connected between a power source and a primary coil, and secondary-side circuitry connected between secondary coil(s) and a load to generate an output voltage and load current for a load. The secondary-side circuitry includes (i) an output inductor, (ii) one or more power transistors, and (iii) control circuitry generating switching signals to operate the power transistors as synchronous rectifiers. The control circuitry (a) senses an inductor current in the output inductor, and (b) selectively disables the switching signals to the power transistors when the inductor current is less than a threshold current value in a range between a minimally negative value and a minimally positive value, the minimally negative value just preventing diode rectification mode of operation of the power transistors, the minimally positive value just preventing negative inductor current in the output inductor.

Full Description:
BACKGROUND 
       [0001]    With the development of power electronics, switch-mode power supplies (SMPS) have become been widely used in many kinds of electronic equipment for their small size, light weight, and high efficiency. A DC/DC power converter is one kind of SMPS used in communication, equipment control and other application fields. 
         [0002]    Saving energy has become important to governments around the world. Electrical devices such as SMPS are rated for their energy efficiency. 
         [0003]    Previously the focus of energy consumption of SMPS was with respect to their efficiency at heavy load or full load, without concern for the power loss at light load or no load . However, there are now standards for power loss at lower loads, such as during standby operation. Thus SMPS are facing the double challenge of decreasing loss at light load or no load and optimizing efficiency at heavy load or full load. 
         [0004]    Another aspect of SMPS is the use of parallel-connected SMPS for higher power applications. In a parallel-connected system, each converter provides only a part of the total output power, so each SMPS can have reduced stress. Also, a set of converters can provide N+M redundancy, so that if one or more parallel-connected converters fail, the overall set can still output 100% of rated power. Other benefits includes the ability to hot swap converters, and the ability to design to different power levels with low design cost. However, connecting a set of SMPS in parallel also presents challenges. If parallel-connected SMPS have different set points, then one SMPS may sink current, and if the sink current is excessive then the SMPS may be damaged or overstressed and thus have a shortened lifetime. 
       SUMMARY 
       [0005]    With the requirements of low power consumption, high efficiency and parallel-connected application, power converters face challenges. The present disclosure is directed to techniques for addressing these challenges. In particular, the disclosure is directed to secondary-side rectification circuitry of a power converter that provides for improved efficiency especially at light load and no load, along with minimization of undesirable current sinking and/or inefficient rectification modes of operating. 
         [0006]    A disclosed power converter employs synchronous rectification mode, providing high efficiency when operating under heavy load or full load. Under light load or no load, the synchronous rectification is modulated to limit or avoid negative output current and thus improve efficiency in these conditions. Modulation occurs when a sensed level of current in an output inductor is less than a predetermined current threshold at or near zero. The exact threshold can be set slightly above or below zero volts in different applications to achieve desired characteristics. By using a slightly positive threshold, negative output current can be avoided. If a small amount of negative output current can be tolerated, then a slightly negative threshold can be used to limit or avoid an inefficient diode rectification mode that can occur by action of a parasitic body diode of the secondary-side power transistors. Overall efficiency in light load and no load conditions can be increased. 
         [0007]    More specifically, a power converter is disclosed that includes a power transformer, primary-side circuitry connected between a power source and a primary coil, and secondary-side circuitry connected between secondary coil(s) and a load to generate an output voltage and load current for a load. The secondary-side circuitry includes (i) an output inductor, (ii) one or more power transistors, and (iii) control circuitry generating switching signals to operate the power transistors as synchronous rectifiers. The control circuitry (a) senses an inductor current in the output inductor, and (b) selectively disables the switching signals to the power transistors when the inductor current is less than a threshold current value in a range between a minimally negative value and a minimally positive value, the minimally negative value just preventing diode rectification mode of operation of the power transistors, the minimally positive value just preventing negative inductor current in the output inductor. The switching signals are effectively modulated in a manner that limits or avoids negative current and/or diode rectification mode, increasing efficiency at light load and no load. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0008]    The foregoing and other objects, features and advantages will be apparent from the following description of particular embodiments of the invention, as illustrated in the accompanying drawings in which like reference characters refer to the same parts throughout the different views. 
           [0009]      FIG. 1  is a schematic diagram of a converter using half-wave diode rectification; 
           [0010]      FIG. 2  is a waveform diagram of signals in the converter of  FIG. 1 ; 
           [0011]      FIGS. 3A and 3B  are schematic diagrams of converters using synchronous rectification; 
           [0012]      FIG. 4  is a schematic diagram of a converter using modulated secondary-side synchronous rectification; 
           [0013]      FIG. 5  is a schematic diagram of analog modulation control circuitry; 
           [0014]      FIGS. 6A and 6B  are schematic diagrams of a pulse width modulator (PWM) generator; 
           [0015]      FIG. 7  is a schematic diagram of a converter using modulated secondary-side synchronous rectification with analog modulation control; 
           [0016]      FIG. 8  is a schematic diagram of digital modulation control circuitry; 
           [0017]      FIG. 9  is a flow diagram of operation of digital modulation control circuitry; 
           [0018]      FIGS. 10A through 13B  are waveform diagrams of operation of converters using modulated secondary-side synchronous rectification under different operating conditions; 
           [0019]      FIGS. 14A through 14C  are schematic diagrams of converters using modulated secondary-side synchronous rectification. 
       
    
    
     DETAILED DESCRIPTION 
       [0020]      FIGS. 1 ,  2 ,  3 A and  3 B illustrate two broad classes or types of rectification used at a secondary side of an isolated converter, diode rectification and synchronous rectification. 
         [0021]      FIG. 1  shows the secondary side of a power converter that provides half-wave diode rectification using a rectifying diode D 1  along with output filtering elements L and C. When the output of the main transformer T 1  is a high level, the rectifier diode D 1  turns on, and the output of diode negative pin (voltage V 1 ) is a high level. When the secondary side output of the main transformer T 1  is a low level, the rectifying diode D 1  turns off, and the output of the diode negative pin is zero.  FIG. 2  shows the waveforms for the transformer secondary voltage VS and the voltage V 1 . In this kind of circuit, rectification conduction loss is proportional to the product of the forward conduction voltage across Dl and the forward conduction current flowing through D 1 . Thus, diode rectification is not optimized for converter efficiency in heavy load or full load. 
         [0022]      FIGS. 3A and 3B  illustrate an alternative rectification technique referred to as “synchronous rectification”. In synchronous rectification, power MOSFETs Q 1 , Q 2  replace the rectification diode(s) at the secondary side of the isolated converter, thus providing high efficiency especially in low-output-voltage and high-current power supplies. Synchronous rectification technology is increasing the DC/DC converter efficiency. A power MOSFET is a voltage controlled component, and when the power MOSFET turns on, the its voltage-ampere characteristic is linear. When using a power MOSFET as a rectifier, the AC gate voltage must be synchronous with the AC rectified voltage, hence the name “synchronous rectifier”. 
         [0023]    Synchronous rectification circuits are shown in  FIGS. 3A and 3B .  FIG. 3A  is a half-wave rectification circuit, and  FIG. 3B  is a full-wave rectification circuit. In both these circuits, Q 1  and Q 2  are power MOSFETs. In the circuit of  FIG. 3A , when the secondary side output voltage of the main transformer T 1  is high level, Q 1  turns on and Q 2  turns off. Q 1  has the function of rectification. When the secondary side output voltage of the main transformer T 1  is low level, Q 1  turns off and Q 2  turns on. Q 2  has the function of free wheeling. The power consumption of the synchronous rectification circuit is mainly the conduction loss and switching loss. When the switching frequency is lower, the conduction loss is the main power loss. When the switching frequency is higher, the switching loss is the main power loss. It is effective to optimize the power loss of MOSFET by selecting a suitable switching frequency. 
         [0024]    Compared with diode rectification, synchronous rectification has lower conduction loss, and it has higher efficiency in heavy load. But in light load or no load, the conduction loss and the switching loss for synchronous rectification are relatively higher. So synchronous rectification mode may be limited in its ability to decrease power loss in light load or no load. 
         [0025]      FIG. 4  shows a converter circuit that uses simulated diode rectification in light load or no load, and synchronous rectification in heavy load or full load, thus decreasing power loss in light load or no load and increasing the efficiency in heavy load or full load. These two modes of operating are referred to herein as “diode rectification mode” and “synchronous rectification mode”. The dividing line between the two modes is based on the level of output current, i.e., the level of current I L  in the inductor L. Usually, when a converter is operating in light load or no load, the output inductor current I L  may become negative. And when the converter is operating in heavy load or full load, the output inductor current I L  is positive. Thus the level of the inductor current I L  can be sensed and used to select the operating mode. 
         [0026]    Specifically, the circuit samples the inductor current I L  , and it compares I L  with a predetermined inductor current switching point I Switch . When I L &gt;I Switch , Q 1  and Q 2  are operated in normal synchronous rectification mode. When I L &lt;I Switch , both Q 1  and Q 2  are disabled (turned off), and the circuit operates in diode rectification mode via parasitic body diodes of Q 1  and Q 2 . Through sensing the inductor current I L , the falling edge of the switching of the secondary side MOSFETs Q 1 , Q 2  is modulated to decrease the root mean square (RMS) value of the inductor current I L , which is effective to lower the loss in light load or no load. 
         [0027]    There are two major ways to realize secondary side MOSFET modulation mode, i.e., an analog way and a digital way. Using the analog way, modulation can be achieved by (1) enabling and disabling a secondary side driver based on output inductor current, or (2) enabling and disabling a secondary side pulse-wide modulator (PWM) controller based on output inductor current. 
         [0028]      FIG. 5  shows a circuit providing secondary side MOSFET modulation by enabling a secondary side driver  12 . Specifically, a comparator  10  compares the inductor current I L  with the threshold current I Switch , and when I L &lt;I Switch , a comparator output signal EN switches to a zero value. This comparator output EN drives an enable pin of a secondary side driver  12  that generates the gate drive signals for the secondary side switching devices (e.g., Q 1 , Q 2  in  FIG. 4 ). The secondary side switching devices turn off synchronized with output inductor current I L . 
         [0029]      FIGS. 6A and 6B  show the secondary side MOSFET modulation mode realized by the peripherals of a PWM controller integrated circuit (PWM generator IC)  20 , which can be either an analog IC or a digital IC. The comparator Comp of the PWM controller IC  20  can reset the output pin of a secondary side PWM to realize modulation of the secondary side MOSFET falling edge from driver  22 .  FIG. 6A  uses an external reference I Switch  for the comparator Comp, while  FIG. 6B  uses an internal reference. 
         [0030]      FIG. 7  illustrates enabling and disabling a secondary side driver  30  based on the level of inductor current I L . When I L &gt;I Switch , the output of a comparator  32  is high. The secondary side driver  30  is enabled, so Q 1  and Q 2  ( FIG. 4 ) operate in normal synchronous rectification mode. When I L &lt;I Switch , the output of the comparator  32  is low. The secondary side driver  30  is disabled, so both Q 1  and Q 2  are disabled. 
         [0031]      FIG. 8  is a block diagram of secondary-side synchronous rectification modulation in a digital way, using a PWM controller IC  40 .  FIG. 9  illustrates operation. A comparator Comp compares the inductor current I L  with the threshold current I Switch , and when I L &lt;I Switch  a comparator interrupt is generated (step  50  in  FIG. 9 ). A comparator interrupt routine Comp Int is then entered (step  52 ), and this routine disables the secondary side PWM for the remaining of the PWM period (step  54 ), similar to the analog way. The interrupt routine is then exited (step  56 ). The threshold I Switch  can be provided by an external reference or an internal reference. 
         [0032]    Compared to the analog way, the digital way to realize the secondary side PWM turning off synchronously may have a little delay. But it can still realize the function of turning off the secondary side PWM synchronously. 
         [0033]      FIGS. 10A-10C ,  11 A- 11 C, and  12 A- 12 C illustrate operation under different loading conditions when different values of the threshold inductor current I Switch  are used. By setting I Switch  appropriately, the secondary side synchronous rectification modulation method can meet different requirements. These figures illustrate a series of cycles of a PWM signal used to control conduction of primary-side transistors (Q PRI ) and second-side transistors (Q SEC ). 
         [0034]      FIGS. 10A ,  10 B and  10 C show operation when I Switch =0. 
         [0035]      FIG. 10A  shows heavy or full load operation in which the minimum of I L  is greater than I Switch . In this case the output inductor current I L  is continuous, and the converter works in continuous conduction mode (CCM). 
         [0036]      FIG. 10B  shows intermediate-load operation in which the minimum of I L  is equal to I Switch . In this case the output inductor current I L  is boundary continuous, and the converter works in the boundary conduction mode (BCM). 
         [0037]      FIG. 10C  shows light or no-load operation in which the minimum of I L  is less than I Switch . In this case the output inductor current I L  is discontinuous, and the converter works in the discontinuous conduction mode (DCM). 
         [0038]      FIGS. 10A-10C  illustrate that selecting I Switch =0 can help avoid current “sinking”, i.e., current flowing into the converter from a load, and thus this selection can be beneficial when converters are arranged in parallel and/or require robust pre-bias startup. Meanwhile, efficiency in full load is not affected. Due to avoiding the negative current, the RMS value of inductor current is decreased, and the power loss is lower in light load and no load. 
         [0039]      FIGS. 11A ,  11 B and  11 C show operation when I Switch &gt;0. 
         [0040]      FIG. 11A  shows that under the condition of the minimum of I L  being greater than or equal to I Switch , the output inductor current I L  is continuous. The converter works in CCM and it is in heavy load. 
         [0041]      FIG. 11B  shows that under the condition of the minimum of I L  being less than I Switch  but nonnegative (I L ≧0), the output inductor current I L  is boundary continuous. During the periods when I L &lt;I Switch , the converter is under diode rectification. The converter works in BCM and it is with intermediate load. 
         [0042]      FIG. 11C  shows that under the condition of the minimum of I L  being less than zero, the output inductor current I L  is discontinuous. During the periods when 0&lt;I L &lt;I Switch , the converter is under diode rectification mode. The converter works in DCM and it is in light load or no load. 
         [0043]    Therefore, selecting I Switch &gt;0 is better to avoid the sink current, thus realizing converters in parallel and pre-bias start well. When I Switch  is close to zero, it does not affect the efficiency in full load. If I Switch  is higher, the converter will have a longer time working under diode rectification mode (0&lt;I L &lt;I Switch ), and thus lower efficiency in heavy load or full load. Otherwise, due to avoiding the negative current, the RMS value of inductor current is decreased, and the power loss is lower in light load and no load. 
         [0044]      FIGS. 12A ,  12 B and  12 C show operation when I Switch &lt;0. 
         [0045]      FIG. 12A  shows that under the condition of the minimum of I L  being greater than or equal to I Switch , the output inductor current I L  is continuous. The converter works in CCM and it is in any load. 
         [0046]      FIG. 12B  shows that under the condition of the minimum of I L  being equal to I Switch , the output inductor current I L  is boundary continuous. The converter works in BCM and it is in any load. 
         [0047]      FIG. 12C  shows that under the condition of the minimum of I L  being less than I Switch , the output inductor current I L  is discontinuous. The converter works in DCM and it is in light load or no load. 
         [0048]    Therefore, selecting I Switch &lt;0 is better to avoid the diode rectification mode. When I Switch  is close to zero, there is little or no sink current, thus realizing converters in parallel and pre-bias start well. When I L  approaches I Switch , the secondary side MOSFETs have overstress issues at the moment of turning off the MOSFETs for the inductor current I L  changing from negative value to zero suddenly. Otherwise, due to the negative current, the RMS value of inductor current I L  is not decreased obviously, and the power loss is still high in light load and no load. 
         [0049]    As shown above, the value of I Switch  should be selected close to 0. Selecting I Switch &lt;0 is better to avoid the diode rectification mode. Selecting I Switch &gt;0 is better to avoid negative current and stress issues. The secondary side synchronous rectification modulation method improves paralleling of converters and pre-bias start. This can be appreciated by comparing  FIGS. 13A and 13B , which show that for conditions in which a first output set point of the converter is lower than the pre-bias voltage there are the different waves between synchronous rectification mode and synchronous rectification modulation mode.  FIG. 13A  shows conventional synchronous rectification mode, and it is clear that the converter has negative current.  FIG. 13B  shows synchronous rectification modulation mode as described herein, and the converter has no negative current. 
         [0050]    The synchronous rectification modulation method of this invention can be used in various isolated DC/DC topologies.  FIGS. 14A ,  14 B and  14 C illustrate use of the secondary side synchronous rectification modulation in different converter topologies.  FIG. 14A  shows a full-bridge topology.  FIG. 14B  shows a half-bridge topology.  FIG. 14C  shows a forward topology. 
         [0051]    While various embodiments of the invention have been particularly shown and described, it will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the spirit and scope of the invention as defined by the appended claims.

Technology Classification (CPC): 8