Patent Abstract:
In a wireless transmitter, a class B or class C amplifier is used as a power amplifier for boosting the power level of a signal to be transmitted. To linearize the power amplifier, a feedback mechanism is included in the transmitter. The feedback mechanism provides for adjustment of a feedback gain which affects the overall gain of the signal, thereby effectively controlling the signal power level. In addition, a phase difference between components of the signal and the feedback versions thereof is corrected to increase the linearity of the power amplifier and stabilize the feedback mechanism.

Full Description:
FIELD OF THE INVENTION 
     The invention relates to communications systems and methods, and more particularly to a system and method for realizing high power efficiency in wireless transmission and effective control of transmission power. 
     BACKGROUND OF THE INVENTION 
     It is well known that with no excitation applied to an amplifier, an operating point, known as a “quiescent point” or “DC operating point,” is defined in an active region of the output characteristics of the amplifier. To provide linear amplification for a radio-frequency (RF) input signal for example, a class A amplifier is normally used, which is typically biased to place the DC operating point far enough from both cutoff and saturation regions in its output characteristics. As a result, the input signal excursions to either side of the operating point do not cause the signal to be cut off or the amplifier to be saturated, thus avoiding introducing distortion into the signal. However, a major drawback of the class A amplifier is its low power efficiency due to a relatively high DC input power required by the amplifier, with respect to its output signal power. 
     A class B amplifier is typically biased to cut off a half-cycle of an input signal, with an output current flow only during the positive half-cycle of the signal. As a result, the amplifier output is significantly distorted, with respect to the input signal. Thus, a class B amplifier is unsuitable for a typical linear operation. However, with respect to a class A amplifier, a class B amplifier affords higher power efficiency as the DC input power to the amplifier is relatively low. 
     A class AB amplifier, on the other hand, is typically biased in such a way that the output current flows for more than half of the cycle of the input signal. As a result, a class AB amplifier behaves like a hybrid between the class A and class B amplifiers. Thus, a class AB amplifier causes a lower distortion than a class B amplifier (but a higher distortion than a class A amplifier) to an input signal at a high power level. At the same time, the class AB amplifier realizes higher power efficiency than a class A amplifier. 
     A class C amplifier is typically biased in such a way that the output current flows less than half of the cycle of the input signal. As a result, the class C amplifier affords the highest power efficiency and, unfortunately, also distortion to an input signal of all of the aforementioned amplifiers. 
     A power amplifier is typically used in a transmitter of a digital wireless communications system, e.g., personal communications service (PCS) system, to boost the power level of a digitally modulated signal for transmission thereof. In wireless communications, the power amplifier is required to afford linear amplification to the signal, without introducing significant distortion thereinto. As such, in the prior art, the power amplifier used in a wireless transmitter typically operates in a class A and/or class AB mode to satisfy the linear amplification requirement. 
     However, for example, a transmitter in a wireless telephone handset is normally powered by a rechargeable battery having a limited capacity. The usable transmission time allowed by the battery before recharging thereof increases with the power efficiency afforded by the power amplifier in the transmitter. Since a long allowable transmission time, and accordingly high amplifier power efficiency, is always desired, the prior art power amplifier operating in a class A and/or class AB mode is deficient in that it affords relatively low power efficiency, compared with a class B or class C amplifier. 
     Nevertheless, it is well known that where a class B or class C amplifier is used, a feedback mechanism may be employed to “linearize” the amplifier to afford substantially linear amplification. 
     SUMMARY OF THE INVENTION 
     It appears that a class B or class C amplifier can be used as the power amplifier in the prior art wireless communications system to increase the power efficiency, in conjunction with the aforementioned feedback mechanism which helps linearize the class B or class C amplifier. However, I have recognized that in such an arrangement, because of the feedback mechanism, a gain control in the prior art wireless communications system which is normally used for controlling the transmission power level can no longer be effectively used for that purpose. Such ineffective power control is particularly disadvantageous in wireless communications as the transmission power requirement frequently changes. 
     In accordance with the invention, a second gain control is incorporated in the above feedback mechanism to impart a selected gain in a feedback version of a transmitted signal, thereby effectively control an overall gain, and thus the power level, of the transmitted signal. A difference between a phase of the transmitted signal and that of the feedback version thereof is also reduced to increase the linearity of the power amplifier and stabilize the feedback mechanism. 
    
    
     BRIEF DESCRIPTION OF THE DRAWING 
     FIG. 1 is a block diagram of a wireless transmitter in accordance with the invention; 
     FIG. 2 illustrates an arrangement for detecting a difference between the phase of a signal and that of a feedback version thereof in the transmitter of FIG. 1; and 
     FIG. 3 is a flow chart depicting the steps of a routine for detecting the phase difference and effecting a phase shift based on the detected difference in the transmitter of FIG.  1 . 
    
    
     Throughout this disclosure, unless otherwise stated, like elements, components or sections in the figures are denoted by the same numeral. 
     DETAILED DESCRIPTION 
     FIG. 1 illustrates wireless transmitter  100  embodying the principles of the invention which may be used in a wireless communications system, e.g., a wireless telephone handset in this instance. In transmitter  100 , baseband processor  105  of conventional design receives an analog signal representing, e.g., voice information from the handset. In response to the received signal, processor  105  generates a bit stream representing the voice information, and processes k bits from the bit stream at a time, where k is an integer greater than one. Specifically, based on each k bit ensemble, processor  105  selects one of 2 k  predetermined symbols differing in their phases in a signal constellation, in accordance with a well known quaternary phase shift keying (QPSK) modulation scheme. The selected symbol corresponding to the k bits is represented by an in-phase (I) signal and a quadrature-phase (Q) signal, which are provided by processor  105  onto leads  109   a  and  111   a  , respectively. 
     Comparator  113 , e.g., a conventional differential amplifier, is used to compare the amplitude of the I signal on lead  109   a  with that of a feedback I signal on lead  109   b . Similarly, comparator  117  is used to compare the amplitude of the Q signal on lead  111   a  with that of a feedback Q signal on lead  111   b . The feedback I and Q signals come from feedback section  150  described below, which is not included in a prior art wireless transmitter using a class A or class AB amplifier as the power amplifier therein. Each of comparators  113  and  117  outputs a signal, known as an “error signal,” representing a difference between the amplitudes of the respective input signals to the comparator. It suffices to know for now that the resulting error signals from comparators  113  and  117  constitute negative feedback to power amplifier  131 , which in this instance is a class B or class C amplifier, to afford relatively high power efficiency, with respect to a class A or class AB amplifier used in the prior art wireless transmitter. 
     In a wireless communication system pursuant to, e.g., the North American cellular or personal communications service (PCS) standards, power amplifier  131  is required to afford substantially linear amplification to a signal to be transmitted, without introducing significant distortion thereinto. Although power amplifier  131  which is a class B or class C amplifier normally affording non-linear amplification, the aforementioned negative feedback in a well known manner causes amplifier  131  to be substantially “linearized,” thereby effectively preventing any transmitted signal distortion attributed to the non-linear amplification which would otherwise adversely affect the wireless communications. As a result, transmitter  100  fully meets the above standards. 
     The error signals from comparators  113  and  117  are respectively filtered by low pass filters (LPFs)  119  and  121  to limit their respective basebands. For example, the cutoff frequency of LPFs  119  and  121  may be about 1 MHz in accordance with the North American cellular or PCS IS- 95  standards. The filtered signals are then fed to feedforward gain control section  125  comprising variable gain amplifiers (VGAs)  125   a  and  125   b . Accordingly, VGAs  125   a  and  125   b  each impart a forward gain to the filtered signals. The specific gains imparted by VGAs  125   a  and  125   b  are determined by controller  159 , and are typically used in the prior art wireless transmitter to adjust its transmission power level to effect power control as required in the wireless communications. 
     Thus, like a controller in the prior art wireless transmitter, controller  159  from time to time receives from a remote base station serving transmitter  100  a signal containing information concerning, among other things, the amount of required transmission power level for current transmission. In a conventional manner, controller  159  compares the required transmission power level with the present level detected by power detector  151  described below. Knowing any difference between the required and present transmission power levels, controller  159  outputs power control signals to accordingly adjust the present level to meet the required level. 
     However, unlike the prior art controller, controller  159  sends the power control signals not only to forward gain control section  125 , but also to feedback gain control section  165  which provides a selected feedback gain in accordance with the invention. This stems from my recognition that because of the aforementioned negative feedback in transmitter  100 , the forward gain alone can no longer be used to effectively adjust the transmission power as required. In fact, if transmitter  100  were devoid of feedback gain control section  165 , i.e., by setting the feedback gain therein to a constant, e.g., one, it can be shown that by virtue of the negative feedback, the power level of the ultimate signal transmitted by transmitter  100  could never exceed a given input power determined by the power level of the I and Q signals, no matter what the forward gain is, thus failing to effect any overall power amplification. 
     In accordance with the invention, section  165  is included to allow effective control of the transmission power. Thus, in response to the aforementioned power control signal from controller  159 , VGA  165   a  and VGA  165   b  in section  165  impart a selected feedback gain to the feedback I and Q signals, respectively, which effectively affects the overall gain of the ultimate, transmitted signal. As a result, the power level of the transmitted signal readily meets the transmission power requirement imposed by the base station. 
     The above amplified signals from VGAs  125   a  and  125   b  are used to respectively modulate two orthogonal carriers having a frequency f c  in modulator  127 . Illustratively, the signal from VGA  125   a  is multiplied by cos(2πf c t) using mulitiplier  127   a  in modulator  127 , where t denotes time. At the same time, the signal from VGA  125   b  is multiplied by sin(2πf c t) using multiplier  127   b . The resulting modulated signals are summed at adder  127   c , resulting in a double sideband carrier signal. The power level of this signal is boosted by power amplifier  131  described above, with its resulting power level meeting the current transmission power requirement. The power-amplified signal is routed by directional coupler  137  to antenna  145  for transmission thereof, thereby realizing wireless communication of the voice information in the transmit direction. 
     At the same time, coupler  137  feeds back an identical version of the transmitted signal to both power detector  151  and attenuator  153  in feedback section  150 . Power detector  151  of conventional design detects the power level of the transmitted signal, and thus the current transmission power level in transmitter  100 . Power detector  151  sends a signal representative of the detected power level to controller  159 , where the received signal is processed to control the transmission power as described before. Attenuator  153  is used to reduce the power level of the transmitted signal by a predetermined factor. The attenuated signal is fed to demodulator  170 . 
     It should be noted at this point that in order to effectively linearize power amplifier  131  and stabilize the feedback loop in feedback section  150 , the phase difference between the I signal on lead  109   a  and the feedback I signal on lead  109   b , and an identical phase difference between the Q signal on lead  111   a  and the feedback Q signal on lead  111   b , need to be controlled. This phase difference, denoted φ, arises from a time lag of each feedback signal behind its counterpart due to the propagation delay (e.g., incurred by the feedback loop) and processing delay (e.g., incurred by power amplifier  131 ) imposed on the feedback signal. 
     Illustratively, the phase difference φ is corrected by shifting the phase of the demodulating carriers used in demodulator  170  by φ, with respect to the modulating carriers used in modulator  127 . As fully described below, demodulator  170  is used to recover a version of the I and Q signals based on the attenuated, transmitted signal from attenuator  153 . Thus, in this instance, the demodulating carriers used in demodulator  170  are cos(2πf c t+φ) and sin(2πf c t+φ), respectively. The actual value of φ is provided by controller  159  to demodulator  170 . This value needs to be initialized when transmitter  100  is powered on, and revised when the required transmission power is changed. 
     In this particular illustrative embodiment, power amplifier  131 , and thus its processing delay which varies with different transmission power required thereof, is fully characterized. Since other delays including the propagation delay is virtually constant, the necessary phase shifts, i.e., φ&#39;s, corresponding to the different transmission power levels can be predetermined, and stored in a memory (not shown) in controller  159 . Thus, when φ needs to be initialized or changed, controller  159  retrieves from the memory the φ value corresponding to the transmission power level requirement from the base station. Controller  159  then sets the feedforward gain of VGAs  125   a  and  125   b , which determines the input power level of power amplifier  131 . At the same time, controller  159  sends the retrieved φ information to demodulator  170  to effect the necessary phase shift. Controller  159  thereafter sets the feedback gain of VGAs  165   a  and  165   b  to achieve the required transmission power level. A second embodiment involving use of a phase detector to determine in real time the amount of the necessary phase shift is described below. 
     In any event, based on the φ value received from controller  159 , demodulator  170  uses multiplier  170   a  to multiply the attenuated, transmitted signal from attenuator  153  and cos(2πf c t+φ) incorporating the received φ value, and multiplier  170   b  to multiply same and sin(2πf c t+φ) also incorporating the received φ value. As mentioned before, the demodulated signal from multiplier  170   a  represents a version of the I signal on lead  109   a . Similarly, the demodulated signal from multiplier  170   b  represents a version of the Q signal on lead  110   a . The demodulated signals are fed to feedback gain control section  165  through leads  173  and  175 , respectively. VGA  165   a  and  165   b  in section  165  respectively impart a selected feedback gain to the demodulated signals, in response to a power control signal from controller  159  as described before. The resulting signals from VGA  165   a  and  165   b  comprise the aforementioned feedback I signal and feedback Q signal, which are provided onto leads  109   b  and  111   b , respectively. 
     The second embodiment involving use of a phase detector for determining in real time the phase difference φ between the Q (or I) signal and the feedback Q(or I) signal will now be described. FIG. 2 shows only the relevant components of transmitter  100  which are in cooperation with the phase detector, denoted  205 , to detect the phase difference. FIG. 3 illustrates routine  300  stored in the memory of controller  159  for detecting such a phase difference, thereby effecting the phase shift based thereon. 
     Referring to both FIGS. 2 and 3, when transmitter  100  is powered on or when the required transmission power needs to be changed, instructed by routine  300 , controller  159  sets the φ value in demodulator  170  to zero, as indicated at step  301 . At step  305 , controller  159  sets the feedback gain of the VGAs in section  165  to zero, thereby opening the feedback loop. Upon learning the current, required transmission power level from the base station, controller  159  at step  310  sets the feedforward gain of the VGAs in section  125  to a selected value, thereby causing power amplifier  131  to deliver the required transmission power level. At step  315 , controller  159  causes phase detector  205  to measure the phase difference between the Q signal on lead  111   a  and the demodulated signal on lead  175  corresponding to the feedback Q signal. Upon learning the detected phase difference from detector  205 , controller  159  at step  320  sets the φ value in demodulator  170  to the detected phase difference value to effect the necessary phase shift. 
     The foregoing merely illustrates the principles of the invention. It will thus be appreciated that a person skilled in the art will be able to devise numerous arrangements which, although not explicitly shown or described herein, embody the principles of the invention and are thus within its spirit and scope. 
     For example, transmitter  100  is disclosed herein in a form in which various transmitter functions are performed by discrete functional blocks. However, any one or more of these functions could equally well be embodied in an arrangement in which the functions of any one or more of those blocks or indeed, all of the functions thereof, are realized, for example, by one or more appropriately programmed processors.

Technology Classification (CPC): 7