Patent Abstract:
A synchronization data detecting unit is provided in a communication system for detecting predetermined synchronization data. The synchronization data are sent in a transmission frame in a communication signal of the communication system. The synchronization data include a sequence of identical binary symbols which are transmitted after scrambling. The synchronization detection unit comprises a descrambler for descrambling the received communication and for producing an output data sequence having multi-level signal values. The multi-level signal values of the descrambler output are smoothed in a filter unit. The smoothed signal is compared with a predetermined threshold value. If the smoothed signal exceeds the predefined threshold value, a detection of said synchronized data is indicated.

Full Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The invention generally relates to a synchronization data detection unit and a method for detecting predetermined synchronization data, and in particular to a receiver and a receiving method in a wireless local area network (WLAN) communication system. 
     2. Description of the Related Art 
     In a communication system such as a wireless local area network (WLAN) system, it is important for a receiver to be synchronized to the transmitter so that messages can successfully be exchanged between the transmitter and the receiver. A wireless local area network system is a flexible data communication system implemented as an extension to or an alternative for a wired LAN. WLAN systems transmit and receive data over the air using radio frequency or infrared technology to minimize the need for wired connections. Thus, WLAN systems combine data connectivity with user mobility. 
     Most WLAN systems use spread spectrum technology, a wide-band radio frequency technique developed for use in a reliable and secure communication system. The spread spectrum technology is designed to trade-off band-width efficiency for reliability, integrity and security. Two types of spread spectrum radio systems are frequently used: frequency hoping and direct sequence systems. 
     In direct sequence spread spectrum systems, spreading is achieved by encoding each data bit using a code word or symbol that has a much higher frequency and information bit rate. The resultant “spreading” of the signal across a wider frequency bandwidth results in a comparatively lower power spectrum density, so that other communication systems are less likely to suffer interference from the device that transmits the direct sequence spread spectrum signal. Direct sequence spread spectrum employs a pseudo random noise code word known to the transmitter and receiver to spread the data. The code word consists of a sequence of “chips” that are multiplied by (or exclusive-ORed) with the information bits to be transmitted. Many wireless networks conform the IEEE 802.11 standard which employs the well-known Barker code word to encode and spread the data. The Barker code word consists of a predefined sequence of eleven chips. One entire Barker code word sequence is transmitted at the time period occupied by an information-containing symbol. 
     To allow higher data rate transmissions, the IEEE 802.11 standard was extended to IEEE 802.11b. In addition to the 11-bit Barker chip, the 802.11b standard uses an 8-bit complementary code keying (CCK) algorithm for high data rate transmission. 
     The data transfer rate may also be improved above the symbol rate by employing higher order modulation techniques, including quadrature phase-shift keying (QPSK) modulation. According to such modulation techniques, each bit is represented by a higher number of possible phases. The transmitter therefore generates two signals, the first signal is called the “in-phase” (I) signal or “I channel” and the second signal is called the “quadrature” (Q) signal or “Q channel” for a 90 degree phase-shifted sinusoidal carrier at the same frequency. 
     The IEEE 802.11 standard for wireless LANs using direct sequence spread spectrum techniques employ a training preamble to train a receiver to a transmitter. Each transmitted data message comprises an initial training preamble followed by a data field. The preamble includes a synchronization field to ensure that the receiver can perform the necessary operations for synchronization. For the preamble length, two options have been defined, namely a long and a short preamble. All compliant 802.11b systems have to support the long preamble. The short preamble option is provided in the standard to improve the efficiency of the network throughput when transmitting special data such as voice or video. The synchronization field of a preamble consists of 128 one bits for a long preamble and 56 zero bits for a short preamble. 
     A receiver detects the synchronization symbols and aligns the receivers internal clock with the symbols in the synchronization field in order to establish a fixed reference time frame with which it interprets the fields in the transmission frame structure following the preamble. The preamble, including the synchronization field, is transmitted with the start of every message (data packet). 
     The purpose of a preamble detection unit is to continuously monitor the incoming signal for the preamble and to indicate if the preamble has been detected. The boundaries between consecutive Barker symbols or CCK symbols are determined and the forwarding of the symbols is to be synchronized to the receiver&#39;s processing schedule. Based on the preamble detection and a timing offset between a symbol arrival and a processing schedule of the following modules, the incoming signal is synchronized to the receivers processing schedule. 
     Referring now to  FIG. 1 , a detection process for detecting a preamble in a communication signal is illustrated. A preamble detection step  101  is performed after receiving a communication signal  100  and before subjecting the received communication signal to further processing, in particular to descrambling  102 . 
     The configuration of a conventional preamble detector  200  is illustrated in FIG.  2 . The received communication signal  201  consisting of an in-phase and a quadrature component is provided to preamble detector  200 . In the preamble detector  200 , the received communication signal  201  is first applied to a despreader  204 , in particular a Barker matched filter (BMF). The despread communication signal is supplied to a demodulator (DEM)  205  for demodulating the despread communication signal. The demodulated signal consists of a sequence of “hard” decisions of the received bit sequence, i.e. each data value of the demodulated signal takes one of both possible binary values. The demodulated bit stream is monitored for detecting the predefined preamble data. Typically, a correlator (e.g. correlator  203 ) is used to detect the preamble. The correlator is essentially a matched filter for the preamble sequence. The correlator produces an output with a large magnitude when the preamble is present. Preamble detection is normally declared when the magnitude of the correlation exceeds a predefined threshold. 
     After preamble detection, the demodulated communication signal is applied to a (digital) descrambler (DDS)  206 . An example of a prior art descrambler  300  (which may be simalar to DDS  206  of  FIG. 2 ) is illustrated in FIG.  3 . The incoming signal  301  a is supplied to delay blocks  304 ,  305  denoting a time delay of several units in accordance with a predefined descrambling rule. The delayed signals are fed back and combined using a multiplicator or exclusive-OR gate  306 . The output is fed back to the incoming signal  301   b  and combined using a multiplier or exclusive-OR gate  303  to produce a descrambled output  302 . 
     Synchronization data detecting units still have a number of problems. One problem is that noise may degrade the signal quality so that the synchronization unit, in particular the preamble detector, fails to declare a preamble even though a preamble is present in the received communication signal. Noise may also produce an output exceeding the threshold when an actual preamble is not present. 
     SUMMARY OF THE INVENTION 
     An improved synchronization detection unit and method are provided that enable a less error prone detection of predefined synchronization data. 
     In one embodiment, a synchronization data detection unit is provided in a communication system for detecting predetermined synchronization data of a transmission frame in a communication signal. The synchronization data include a sequence of identical binary symbols which are transmitted after being sent through a scrambler. The synchronization data detection unit comprises a descrambler for descrambling the received communication signal and for producing an output data sequence having multi-level signal values. The multi-level signal values are applied to a filter means for smoothing the descrambler output. The smoothed signal is supplied to a threshold means. The threshold means compares the smoothed output of the filter means with a predetermined threshold value. If the output of the filter means exceeds the predefined threshold value, the threshold means indicates the detection of said synchronization data. 
     In another embodiment, a synchronization data detection unit is provided in a communication system for detecting predetermined synchronization data of a transmission frame in a communication signal. The synchronization data include a sequence of identical binary symbols which are transmitted after being sent through a scrambler. The synchronization data detection unit comprises a descrambler for descrambling the received communication signal and for producing an output data sequence having multi-level signal values. The multi-level signal values are applied to a filter for smoothing the descrambler output. The smoothed signal is supplied to a comparator. The comparator compares the smoothed output of the filter with a predetermined threshold value. If the output of the filter exceeds the predefined threshold value, the comparator indicates the detection of said synchronization data. 
     In still another embodiment, a method for receiving synchronization data is provided for use in a communication system. The predetermined synchronization data are included in a transmission frame of a communication signal. The synchronization data comprise a sequence of identical binary symbols which are transmitted after scrambling. The received communication signal is descrambled to produce a sequence of multi-level output signal values. The multi-level output signal values are smoothed and the smoothed signal is compared to a predetermined threshold value. If the smoothed signal exceeds the predetermined threshold, said synchronization data are detected. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The accompanying drawings are incorporated into and form a part of the specification for the purpose of explaining the principles of the invention. The drawings are not to be construed as limiting the invention to only the illustrated and described examples of how the invention can be made and used. Further features and advantages will become apparent from the following and more particular description of the invention, as illustrated in the accompanying drawings, wherein: 
         FIG. 1  is a flow-chart illustrating a preamble detection procedure; 
         FIG. 2  is a block diagram illustrating a preamble detector for detecting a preamble in a communication signal; 
         FIG. 3  is a block diagram illustrating the configuration of a descrambler incorporated into a preamble detector as shown in  FIG. 2 ; 
         FIG. 4  is a block diagram illustrating a synchronization data detection unit; 
         FIG. 5  is a block diagram illustrating a configuration of a half-soft descrambling module incorporated in the configuration as shown in  FIG. 4 ; 
         FIG. 6  is a block diagram illustrating a more detailed embodiment of the descrambling module shown in  FIG. 5 ; 
         FIG. 7  is a block diagram illustrating a configuration of a filter module as shown in the configuration of  FIG. 4 ; 
         FIG. 8  is a block diagram illustrating another configuration of a filter module as shown in  FIG. 6 ; 
         FIG. 9  is a flow-chart illustrating a preamble detection processing procedure; 
         FIG. 10  is a flow-chart illustrating a threshold comparison procedure for detecting predetermined synchronization data in a communication signal; and 
         FIG. 11  is a flow-chart illustrating a more detailed threshold comparison procedure for detecting two different kinds of preambles in a communication signal. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     The illustrative embodiments of the present invention will be described with reference to the figure drawings. 
     Referring now to the drawings and in particular to  FIG. 4 , which illustrates a synchronization data detecting unit for detecting synchronization data, in particular a predefined preamble, as described herewith. The configuration as shown in  FIG. 4  consists of a synchronization data detecting module  400  and a synchronizing module  403 . The synchronization data detecting module  400  detects a preamble and may also provide a timing offset between a symbol arrival and the processing schedule of the following modules. The symbol synchronizer  403  will use the timing offset information to synchronize the data stream to the processing schedule. 
     The synchronization data detecting module  400  comprises the following modules: a Barker matched filter (BMF) module  404 , a differential BPSK demodulator (DEM) module  405 , and a soft descrambler (SDS) module  406 . These modules, i.e. BMF, DEM, and SDS, form together a non-coherent receiver. Further, the synchronization data detecting module  400  comprises a comb filter (COF) module  407  for smoothing the descrambler output. For evaluating the smoothed data to detect the predefined preamble data the synchronization data detecting module  400  contains a threshold controller  408 . Details of the above-identified modules are described below. 
     The Barker matched filter module  404  receives the communication signal  401  input to the synchronization data detecting module  400  and computes a correlation between the Barker sequence and the samples of the input signal. The Barker code word consists of eleven chips which comprise the sequence “01001000111” or “+1,−1,+1,+1,−1,+1,+1,+1,−1,−1,−1,” (non-return-zero-NRZ), in which the leftmost chip is output first in time. One entire Barker code word sequence is received in the time period occupied by an information-containing symbol. Thus, if the symbol rate is 1 Mbaud, the underlying chip rate for the eleven chips of the Barker sequence is 11 MHz. By using the 11 MHz chip rate signal, the spectrum occupied by the transmitting signal is eleven times greater. In an exemplary implementation, the Barker sequence is extended from eleven samples to twenty-two samples due to an input sample rate of 22 Msps. This is achieved by stuffing zeros between the original elements of the Barker sequence. 
     The input samples of the I channel and the Q channel are correlated with the Barker sequence, respectively. A complex correlation sample is computed for each complex input sample. This may be implemented by a “sliding-window” algorithm known in the art. 
     The despread communication signal is applied to demodulator module  405 . In a specific implementation, the demodulator module  405  is a differential BPSK demodulator. 
     Turning now to FIG.  5  and  FIG. 6 , which illustrate particular implementations for a descrambler module  406 . The descrambler module  406  may be implemented as a soft-descrambler outputting multi-value samples. In contrast to digital demodulators, in a soft descrambler multi-value input signals are not subjected to a “hard” binarization, but the descrambled output values are aligned around the expected binary values. 
     In other embodiments described herein, a “half-soft” descrambler is used for descrambling the demodulated signal. Examples of a configuration of a “half-soft” descrambler module  500  are shown in FIG.  5  and  FIG. 6. A  half-soft descrambler differs from a soft descrambler in that the input signal  501   a  applied to a “delay portion”  504 - 506  in said descrambler is binarized whereas the other branch of the input signal  501   b  is maintained as multi-value signal. In descrambler  500  of  FIG. 5 , the received input signal  501   a  is applied to a binarization portion  503  for converting a multi-value input sample value into a binary value. 
     The descrambler  500  is configured in accordance with a predefined generating polynomial, e.g. given by 1+x a +x b  wherein x a  and x b  denote predefined time delays. The delayed signals are fed back and added to the input signal using a multiplication or exclusive-OR gate to produce the descrambled output. As shown in  FIG. 6 , the delay elements  504 ,  505  comprise a plurality of one bit registers  601 - 606 , each denoting a time delay of one unit. The half-soft descrambler outputs a sequence of soft-symbols aligned around the expected binary values. 
     The half-soft descrambler arrangement of FIG.  5  and  FIG. 6  may have the advantage that the hardware effort for providing multi-level output samples is considerably reduced compared to conventional soft descramblers. 
     When receiving preamble data, the multi-level output sample values of a soft or half-soft descrambler all have the same sign. In order to reduce an influence of random deviations in the output sample values, the output sample valules are averaged by means of a comb filter  407 . Implementation examples for a comb filter are shown in FIG.  7  and FIG.  8 . 
     Referring now to  FIG. 7 , the incoming real and imaginary data symbols  701  are applied to a multiplier  705  to be weighted with a weight W 1 . The weighed signal is added to a fed back and delayed output signal  702  using a delay element  704  and an adder  703 . Before adding the fed back output signal and the input signal, the delayed fed back signal is also weighted using a multiplier  706  with a weight W 2 . 
     According to a particular embodiment, the incoming data symbols  701  are multiplied in multiplier  705  with a fixed-point equivalent of 0.2 and the delayed “averaged” values provided by delay element  704  are multiplied with a fixed-point equivalent of 0.8. Although the algorithm of this embodiment is described with reference to weighting values of 0.2 and 0.8, respectively, those skilled in the art will appreciate that the weighting algorithm may be implemented to the same effect using a variety of weighting value combinations W 1  and W 2 . A noise reduction of the input sample values  701  may be increased by reducing the amount of weight W 1  and increasing the amount of weight W 2 . In contrast, the accuracy of the received data symbols in time may be increased by increasing the amount of the first weight W 1  and decreasing the amount of the second weight W 2 . 
     Another embodiment of a comb filter configuration is shown in FIG.  8 . Each output of the depicted comb filter represents an average of n chip samples that are spaced by eleven or twenty-two samples (depending on the employed input sample rate). As those skilled in the art will appreciate, the number of averaged samples is set to be appropriate for a sufficient noise reduction. Noise will be reduced more efficiently when increasing the number n. In a specific example of this embodiment, the number n of averaged samples is 10. An averaged amplitude is computed separately for the in-phase channel I and the quadature channel Q. This may be achieved in the specific embodiment by implementing the following formulas: 
                 f   I     ⁡     (   k   )       =       1   10     ⁢       ∑     i   =   0     9     ⁢       d   I     ⁡     (     k   -     22   ⁢   i       )                           f   Q     ⁡     (   k   )       =       1   10     ⁢       ∑     i   =   0     9     ⁢       d   Q     ⁡     (     k   -     22   ⁢   i       )                     
 
wherein f(k) represents the comb filter output and d(k−22i) represents the decrambler output when employing a Barker sequence of 22 samples, as mentioned above.
 
     Both filter outputs f I  and f Q  will be used to decide if a preamble is currently received. In a specific embodiment, both outputs are added to counter the influence of the frequency offset of the signal:
 
 S ( kT )= f   I ( kT )+ f   Q ( kT )
 
wherein S represents the sum of both outputs and f I  and f Q  represent the averaged comb filter output of the respective channel.
 
     Referring back to  FIG. 8 , comb filter  800  comprises a plurality of delay elements  804 , an adder  803  and a divider  805 . The incoming descrambler output  801  is applied to the adder  803  and branched to a first one of the identical delay elements  804 . The output of each of the delay elements  804  is applied to adder  803  and to the input of a subsequent one of the delay elements  804 . Adder  803  receives a predetermined number n of input signals which are accumulated and provided to divider  805 . The number n of inputs and a corresponding number of n−1 of delay elements  804  is set in accordance with the above described considerations. Divider  805  normalizes the accumulated sum provided by adder  803 . 
       FIG. 9  schematically shows an example of how to detect a preamble in an incoming communication signal. After receiving (step  900 ) the communication signal, the communication signal is subjected to a soft descrambling process  901  and the descrambled output is smoothed in order to reduce the influence of random deviations in the communication signal ( 902 ). The preamble is detected in the output signal on the basis of the output signal values of the averaged descrambler output in step  903 . The preamble detection is performed by comparing the comb filter output against a predefined threshold value. Details of the process performed by threshold controller  408  are illustrated in FIG.  10 . 
     As shown in  FIG. 10 , each obtained comb filter sum S (step  1001 ) is compared during preamble search against a predefined threshold Th in step  1002 . When the comb filter output exceeds the predefined threshold Th, a preamble may have been detected (step  1003 ) and the preamble detection procedure enters a “preamble detecting state”. Threshold controller  408  will remain in this state while fetching two more chips to determine if subsequent descrambler outputs are even larger. 
     After comparing these two further sample values, the threshold controller will periodically confirm the comparison result, i.e. to determine whether or not a descrambler peek re-appears after a symbol duration. The comb filter output has to exceed the predefined threshold Th a predetermined number of times before a preamble detection is definitely declared (steps  1004 ,  1005 ). The “locked state” (in case of a preamble detection) is entered after the comparator result in step  1002  is confirmed a predetermined number of times T N . In one implementation, T N  is 15, i.e. requiring to confirm a preamble detection 15 times. Those skilled in the art will appreciate that any other number of times may be implemented to the same effect, e.g. a number T N  between 10 and 20 times. 
     In accordance with the IEEE 802.11b standard for wireless LANs, the threshold controller  408  may be adapted to detect a long and a short preamble. As both preambles not only differ by their lengths but also differ by the binary value of the preamble sequence, these preambles may be distinguished by employing different threshold values corresponding the binary preamble values. 
     A threshold comparison process able to detect a long and a short preamble in accordance with the wireless LAN standard is illustrated in FIG.  11 . The preamble detection procedure employs a first threshold Th 1 , and a second threshold Th 2 . In accordance with the expected descrambler output values +1/−1 both threshold values only differ by the sign. 
     After having first detected a long or short preamble in step  1102  or in step  1107 , a “reliability check” is performed for each of the preambles repeatedly either by steps  1103 - 1106  or by steps  1108 - 1111 . In case the detected long or short preamble cannot be confirmed in one of the confirmation loops formed by steps  1103 - 1106  for a long preamble confirmation or by steps  1108 - 1111  for a short preamble confirmation, the threshold comparator returns to step  1101 . 
     After a preamble has been detected and confirmed and a timing offset between symbol arrival and symbol processing in the following data modules is known, the symbol synchronizer module  403  will release the data such that symbol release and symbol processing are matched. 
     According to the various embodiments described above, the decision for each sent symbol is shifted from the demodulator&#39;s output to the comb filter&#39;s output. According to a specific embodiment, a half-soft descrambler is employed which introduces a binarization in the descrambler “delay path” comprising a plurality of one bit width registers. The output of the half-soft descrambler is a sequence of soft symbols which are aligned around the expected value of +1 (long preamble consisting of sent ones) or −1 (short preamble consisting of sent zeros) based on the sent preamble sequence. 
     The embodiments described above may provide a more reliable and improved preamble detection without increasing the hardware complexity. By shifting the preamble detecting step to a subsequent processing stage, the preamble detection of the various embodiments described herein reduce the occurrence of failure to detect a preamble or to wrongly detect a preamble.

Technology Classification (CPC): 7