Patent Abstract:
A multiphase ripple voltage regulator generator employs a hysteretic comparator referenced to upper and lower voltage thresholds. The hysteretic comparator monitors a master ripple voltage waveform developed across a capacitor supplied with a current proportional to the difference between the output voltage and either the input voltage or ground. The output of the hysteretic comparator generates a master clock signal that is sequentially coupled to PWM latches, the states of which define the durations of respective components of the synthesized ripple voltage. A respective PWM latch has a first state initiated by a selected master clock signal and terminated by an associated phase voltage comparator that monitors a respective phase node voltage.

Full Description:
CROSS-REFERENCE TO RELATED APPLICATION  
       [0001]    The present application is a continuation-in-part of co-pending U.S. patent application Ser. No. 10/236,787, filed Sep. 6, 2002, entitled: “Synthetic Ripple Regulator,” by M. Walters et al (hereinafter referred to as the &#39;787 patent application), assigned to the assignee of the present application and the disclosure of which is incorporated herein. 
     
    
     
       FIELD OF THE INVENTION  
         [0002]    The present invention relates in general to power supply circuits and components therefor, and is particularly directed to an arrangement for synchronizing a plurality of synthetic ripple generators that generate artificial or synthesized ripple waveforms to control switching operations of a multiphase DC-DC converter.  
         BACKGROUND OF THE INVENTION  
         [0003]    As described in the background section of the above-referenced &#39;787 patent application, electrical power for integrated circuits is typically supplied by one or more direct current (DC) power sources. In a number of applications the circuit may require plural regulated voltages that are different from the available supply voltage, which may be relatively low e.g., on the order of three volts or less, particularly where low current consumption is desirable, such as in portable, battery-powered devices. (This architecture may achieve a much high voltage difference in portable applications, for example an input voltage on the order of 4.5-25V and an output voltage Vo on the order of 0.5V-3.7V.) Moreover, in many applications the load current may vary over several orders of magnitude. To address these requirements it has been common practice to employ ripple generator-based converters, such as a hysteresis or ‘bang-bang’ converter of the type shown in FIG. 1.  
           [0004]    Such a ripple regulator-based DC-DC voltage converter employs a relatively simple control mechanism and provides a fast response to a load transient. The switching frequency of the ripple voltage regulator is asynchronous, which is advantageous in applications where direct control of the switching frequency or the switching edges is desired. For this purpose, the ripple voltage regulator of FIG. 1 employs a hysteresis comparator  10 , that switchably controls a gate drive circuit  20 , respective output drive ports  22  and  23  of which are coupled to the control or gate drive inputs of a pair of electronic power switching devices, respectively shown as an upper P-MOSFET (or PFET) device  30  and a lower N-MOSFET (or NFET) device  40 . These FET switching devices have their drain-source paths coupled in series between first and second reference voltages (Vdd and ground (GND)).  
           [0005]    The gate drive circuit  20  controllably switches or turns the two switching devices  30  and  40  on and off, in accordance with a pulse width modulation (PWM) switching waveform (such as that shown at PWM in the timing diagram of FIG. 2) supplied by comparator  10 . The upper PFET device  30  is turned on and off by an upper gate switching signal UG applied by the gate driver  20  to the gate of the PFET device  20 , and the lower NFET device  30  is turned on and off by a lower gate switching signal LG applied by the gate driver  20  to the gate of the NFET device  30 .  
           [0006]    A common or phase voltage node  35  between the two power FETs  30 / 40  is coupled through an inductor  50  to a capacitor  60 , which is referenced to a prescribed potential (e.g., ground (GND)). The connection  55  between the inductor  50  and the capacitor  60  serves as an output node, from which an output voltage Vout (shown as triangular waveform Output in FIG. 2) is derived. In order to regulate the output voltage relative to a prescribed reference voltage, the output node  55  is coupled to a first, inverting (−) input  11  of the hysteresis comparator  10 , a second, non-inverting (+) input  12  of which is coupled to receive a DC Reference voltage.  
           [0007]    In such a hysteretic regulator, the output PWM signal waveform produced by hysteresis comparator  10  transitions to a first state (e.g., goes high) when the output voltage Vout at node  55  falls below the reference voltage Reference (minus the comparator&#39;s inherent hysteresis voltage Δ). Conversely, the comparator&#39;s PWM output transitions to a second state (e.g., goes low) when the output voltage Vout exceeds the reference voltage plus the hysteresis voltage Δ. The application of or increase in load will cause the output voltage (Vout) to decrease below the reference voltage, in response to which comparator  10  triggers the gate drive to turn on the upper switching device  30 . Because the converter is asynchronous, the gate drive control signal does not wait for a synchronizing clock, as is common in most fixed frequency PWM control schemes.  
           [0008]    Principal concerns with this type of ripple voltage regulator include large ripple voltage, DC voltage accuracy, and switching frequency. Since the hysteretic comparator  10  directly sets the magnitude of the ripple voltage Vout, employing a smaller hysteresis Δ will reduce the power conversion efficiency, as switching frequency increases with smaller hysteresis. In order to control the DC output voltage, which is a function of the ripple wave shape, the peak  71  and the valley  72  of the output ripple voltage (Output, shown in FIG. 2) is regulated. For the triangular wave shape shown, the DC value of the output voltage is a function of the PWM duty factor. The output voltage wave shape also changes at light loads, when current through the inductor  50  becomes discontinuous, producing relatively short ‘spikes’ between which are relatively long periods of low voltage, as shown by the DISCON waveshape in FIG. 2. Since the ripple voltage wave shape varies with input line and load conditions, maintaining tight DC regulation is difficult.  
           [0009]    In addition, improvements in capacitor technology will change the ripple wave shape. In particular, the current state of ceramic capacitor technology has enabled the equivalent series resistance or ESR (which produces the piecewise linear or triangular wave shape of the output voltage waveform shown in FIG. 2) of ceramic capacitors to be reduced to very low values. At very low values of ESR, however, the output voltage&#39;s ripple shape changes from triangular to a non-linear shape (e.g., parabolic and sinusoidal). This causes the output voltage to overshoot the hysteretic threshold, and results in higher peak-to-peak ripple. As a result, the very improvements that were intended to lower the output voltage ripple in DC-DC converters can actually cause increased ripple when used in a ripple voltage regulator.  
           [0010]    In accordance with the invention disclosed in the &#39;787 application, shortcomings of conventional ripple voltage regulators, including those described above, are effectively obviated by the synthetic ripple voltage regulator shown in FIG. 3. This synthetic ripple voltage regulator generates an auxiliary voltage waveform, that effectively replicates or mirrors the waveform ripple current through the output inductor  50 , and uses this auxiliary voltage waveform to control toggling of the hysteretic comparator  10 . Using such a reconstructed current for the purpose of ripple voltage regulation results in low output ripple and simplified compensation.  
           [0011]    More particularly, the synthetic ripple voltage regulator of FIG. 3 employs a transconductance amplifier  110 , the output of which is coupled to a ‘ripple voltage’ capacitor  120 . The transconductance amplifier  110  produces an output current I RAMP  proportional to the voltage across inductor  50 , which is interconnected between a node  35  common with the upper and lower MOSFETs (respective gate drives  21  and  22  for which are produced by a gate drive circuit  20 ), and an output node  55 . The ripple voltage capacitor  120  transforms this output current ramp into an inductor current-representative voltage having the desired waveform shape. A benefit of synthesizing the ripple waveform based on inductor current is the inherent feed-forward characteristic. For a step input voltage change, the current I RAMP  produced by the transconductance amplifier  110  will change proportionally to modify the conduction interval of the power switching devices.  
           [0012]    For this purpose, transconductance amplifier  110  has a first, non-inverting (+) input  111  coupled to the phase node  35  and a second, inverting (−) input  112  coupled to output voltage node  55  at the other end of inductor  50 , so that the transconductance amplifier  110  effectively ‘sees’ the voltage across inductor  50 . The output voltage node  55  is further coupled to a first terminal  121  of capacitor  120  and to the inverting (−) input  141  of an error amplifier  130  inserted upstream of the hysteresis comparator  10 . Error amplifier  130  serves to increase the DC regulation accuracy, providing high DC gain to reduce errors due to ripple wave shape, various offsets, and other errors. Error amplifier  130  has a second, non-inverting (+) input  132  thereof coupled to receive the voltage Reference, while its output  133  is coupled to the non-inverting (+) input  12  of hysteresis comparator  10 . In the configuration of FIG. 3, the output of the error amplifier  130  follows the load current. The transconductance amplifier  110  has its output  113  coupled to a second terminal  122  of the capacitor  120  and to inverting (−) input  11  of the hysteresis comparator  10 .  
           [0013]    The operation of the synthetic ripple voltage regulator of FIG. 3 may be understood with reference to the set of waveform timing diagrams of FIG. 4. As a non-limiting example, the regulator voltage may be set at a value of Reference=1 VDC and the hysteresis comparator  10  may trip with +/−100 mV of hysteresis. The inductance of inductor  50  is 1 μH and the output capacitance is 10 μF. The line M 1  (at the 30 μsec time mark) in FIG. 4 represents a change in input voltage from a value on the order of 3.6 VDC prior to M 1  to a value on the order of 4.2 VDC at M 1  and thereafter.  
           [0014]    The upper waveform  501  corresponds to the ripple voltage generated across the ripple voltage capacitor  120 ; the middle waveform  502  is the current through inductor  50 , and the lower waveform  503  is the output voltage at node  55 . The similarity of the respective ripple and inductor current waveforms  501  and  502  is readily apparent, as shown by respective step transitions  511 / 521  and  512 / 522  therein, at t=20 μs and t=50 μs. As shown by waveform  502 , the converter is initially supplying an inductor current on the order 100 mA for an input supply voltage of 3.6 VDC. This inductor current is discontinuous and the switching frequency has a relatively stable value on the order of 900 kHz.  
           [0015]    At the transient  521  (t=20 μs) in waveform  502 , there is a stepwise (X10) increase in the load current from 100 mA to a value on the order of 1 A, and the switching frequency increases to a frequency on the order of 1.5 MHz. From the output voltage waveform  503 , it can be seen that the amount of ripple  531  occurring at this transient is relatively small (on the order of only +/−3 mV, which is well below that (+/−100 mV) of the prior art regulator of FIG. 1, during discontinuous operation, where load current=100 mA, and then drops to +/−1.5 mV).  
           [0016]    At the M 1  or t=30 μs time mark, there is a stepwise increase in input voltage from 3.6 VDC to 4.2 VDC, and the switching frequency increases to almost 2.3 MHz, yet the levels of each of waveforms  501 ,  502  and  503  remain stable. Subsequently, at t=50 μs, there is a step transient  512  in the inductor/load current waveform  501 , which drops back down from 1 A to 100 mA, and the switching frequency settles to a value on the order of 1.3 MHz. As can be seen in the output voltage waveform  503 , like the ripple  531  occurring at the t=20 μs transient, the amount of ripple  532  for this further transient is also relatively small (on the order of only +−3 mV and dropping to +/−1.5 mV), so that the output voltage may be effectively regulated at a value on the order of the voltage Reference of 1 VDC.  
         SUMMARY OF THE INVENTION  
         [0017]    In accordance with the present invention, the functionality of the transconductance amplifier and hysteretic comparator architecture disclosed in the &#39;787 application is applied to a multiphase DC-DC voltage generator, to realize a new and improved circuit arrangement for synchronizing a plurality of synthetic ripple voltage generators, that generate artificial or synthesized ripple voltage waveforms for controlling switching operations of a multiphase DC-DC voltage converter. The synthetic ripple voltage regulator of the invention has a variable frequency that is a function of the input voltage, output voltage and load.  
           [0018]    For this purpose, the invention comprises a master hysteretic comparator that is referenced to upper and lower voltage thresholds. The master hysteretic comparator monitors a master ripple voltage waveform that is produced across a capacitor by a current proportional to the difference between the output voltage and either the input voltage or a reference voltage (ground). The proportionality current is produced by a transconductance amplifier pair. The output of the master hysteretic comparator serves as a master clock signal that is sequentially coupled to PWM latches, the states of which define the durations of respective components of the synthesized ripple voltage. A respective PWM latch has a first state thereof initiated by a selected master clock signal produced by the hysteretic comparator and terminated by an associated comparator that monitors a respective phase node voltage.  
           [0019]    As noted above, the synthetic ripple voltage regulator of the invention has a variable frequency that is a function of the input voltage, output voltage and load. In accordance with an alternative approach, a comparator and one-shot are used to generated a master clock signal having a fixed, steady-state frequency, with the difference between Vlower and Vupper being set proportional to the output voltage Vo. In an alternative methodology for producing produce the output signal PWM1, the output signal from the sequence logic causes the output port signal PWM1 to change state (e.g., go high), and a switch is turned on. The ripple capacitor voltage across a ripple capacitor is thereby increased by a charge current proportional to (Vin−Vo). The phase1 ripple voltage crosses the upper voltage threshold Vupper, and a comparator resets the output flip-flop from which PWM1 is produced. This causes the PWM1 output to change state (go low). During the interval between opposite peaks in the phase1 ripple capacitor voltage, the voltage across the capacitor decreases by a discharge current proportional to Vo. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0020]    [0020]FIG. 1 diagrammatically illustrates the general architecture of a conventional ripple regulator-based DC-DC voltage converter;  
         [0021]    [0021]FIG. 2 is a timing diagram showing PWM and output voltage waveforms associated with the operation of the ripple regulator-based DC-DC voltage converter of FIG. 1;  
         [0022]    [0022]FIG. 3 diagrammatically illustrates an implementation of the synthetic ripple voltage regulator in accordance with the invention disclosed in the &#39;787 application;  
         [0023]    [0023]FIG. 4 is a timing diagram showing waveforms associated with the operation of the synthetic ripple voltage regulator of FIG. 3;  
         [0024]    [0024]FIG. 5 diagrammatically illustrates a multiphase synthetic ripple voltage regulator in accordance with the present invention;  
         [0025]    [0025]FIG. 6 contains a set of timing diagrams associated with the operation of the multiphase synthetic ripple voltage regulator of FIG. 5.  
         [0026]    [0026]FIG. 7 shows the use of a single comparator and one-shot to generate a master clock signal;  
         [0027]    [0027]FIG. 8 is a timing diagram associated with the operation of FIG. 7;  
         [0028]    [0028]FIG. 9 illustrates an alternative circuit arrangement for producing an output signal PWM1;  
         [0029]    [0029]FIG. 10 is a timing diagram associated with the operation of FIG. 8;  
         [0030]    [0030]FIG. 11 is a timing diagram of upper and lower voltages associated with a load step;  
         [0031]    [0031]FIG. 12 shows a master clock pulse train associated with the transient increase of FIG. 11; and  
         [0032]    [0032]FIG. 13 graphically illustrates the change in frequency between a first relatively steady state, followed by a transition to a higher frequency and then a return to a further steady state frequency. 
     
    
     DETAILED DESCRIPTION  
       [0033]    Before describing a non-limiting, but preferred embodiment of the multiphase synthetic ripple voltage regulator synchronization scheme of the present invention, it should be observed that the invention resides primarily in an arrangement of conventional circuit components, and the manner in which they may be incorporated into a multiphase hysteretic controller of the type described above. It is to be understood that the invention may be embodied in a variety of other implementations, and should not be construed as being limited to only the embodiment shown and described herein. Rather, the implementation example shown and described here is intended to supply only those specifics that are pertinent to the present invention, so as not to obscure the disclosure with details that are readily apparent to one skilled in the art having the benefit of present description. Throughout the text and drawings like numbers refer to like parts.  
         [0034]    Attention is now directed to FIG. 5, which diagrammatically illustrates the general architecture of a multiphase synthetic ripple voltage regulator in accordance the present invention for a two phase regulator. It will be readily appreciated from the description to follow that the architecture and functionality of the present invention may be readily expanded to additional phases as desired. A two phase implementation has been shown as a reduced complexity multiphase example for purposes of reducing the complexity of the drawings and their attendant description.  
         [0035]    The multiphase synthetic ripple voltage regulator of FIG. 5 is shown as comprising a ‘master’ hysteretic comparator  200  formed of upper and lower threshold comparators  210  and  220 , outputs of which are respectively coupled to the SET and RESET inputs of a SET/RESET flip-flop  230 . A first, inverting (−) input  211  of comparator  210  is coupled to receive an upper threshold voltage Vupper, while first, non-inverting (+) input  221  of comparator  220  is coupled to receive a lower threshold voltage Vlower, that is some prescribed offset ΔV/2 lower than the upper threshold voltage Vupper. Each of the second, non-inverting input  212  of comparator  210  and the second, inverting (−) input  222  of comparator  220  are coupled to a common terminal  241  of a controlled switch  240 , and also to a capacitor  245 , which is referenced to ground. Switch  240  is controlled by the Q output of flip-flop  230 .  
         [0036]    A first input terminal  242  of switch  240  is coupled to the output of a transconductance amplifier  250 , while a second input terminal  243  of switch  240  is coupled to the output of a transconductance amplifier  260 . Transconductance amplifier  250  has a first, non-inverting (+) input  251  coupled to receive the input voltage Vin to the regulator, while a second, inverting (−) input  252  thereof is coupled to receive the output voltage Vo of the regulator (namely, the voltage at output node  55  of the circuits of FIGS. 1 and 3, for example). Transconductance amplifier  250  produces an output current proportional to the difference between its inputs, namely proportional to Vin−Vo. Transconductance amplifier  260  has a first, non-inverting (+) input  261  coupled to ground, while a second input  262  thereof is coupled to receive the output voltage Vo. Transconductance amplifier  250  produces an output current proportional to the difference between its inputs, namely proportional to 0−Vo.  
         [0037]    The QBAR output of flip-flop  230  is coupled to a sequence logic circuit  270 . Sequence logic circuit  270 , which may be implemented as a counter, has N outputs corresponding to the number of phases being generated. In the present two phase example, sequence logic circuit  270  has a first output  271  coupled to the SET input of a SET/RESET flip-flop  280  and a second output  272  coupled to the SET input of SET/RESET flip-flop  290 . For this purpose, sequence logic  270  may be implemented as a flip-flop for a two-phase application, or a shift register in more than a two-phase application. The RESET input of flip-flop  280  is coupled to the output of a comparator  300 , while the RESET input of flip-flop  290  is coupled to the output of a comparator  310 .  
         [0038]    Comparators  300  and  310  have first, non-inverting (+) inputs  301  and  311  respectively coupled to receive the upper threshold voltage Vupper. The inverting (−) input  302  of comparator  300  is coupled to receive a phase 1 ripple voltage waveform that is developed across a capacitor  305 , as a result of current supplied to capacitor  305  by a phase 1 transconductance amplifier  320 . The inverting (−) input  312  of comparator  310  is coupled to receive a phase 2 ripple voltage that is developed across a capacitor  315 , as a result of current supplied to capacitor  315  by a phase 2 transconductance amplifier  330 .  
         [0039]    Phase 1 transconductance amplifier  320  has a first, non-inverting (+) input  321  coupled to receive a phase 1 voltage Vphase1 and a second, inverting (−) input  322  coupled to receive the output voltage Vo. The phase 1 voltage Vphase1 corresponds to the voltage at node  35  of the converter circuit associated with a first phase output voltage, and controllably gated in accordance with the PWM1 waveform output of output flip-flop  280 . Thus, transconductance amplifier  320  generates a voltage Phase1 ripple proportional to Vphase1−Vo. Similarly, phase 2 transconductance amplifier  330  has a first, non-inverting (+) input  331  coupled to receive a phase 2 voltage Vphase2, and a second, inverting (−) input  332  coupled to receive the output voltage Vo. The phase 2 voltage Vphase2 corresponds to the voltage at node  35  of the converter circuit associated with a second phase output voltage, and controllably gated in accordance with the PWM2 output of output flip-flop  290 . Thus, transconductance amplifier  330  generates a voltage Phase2 ripple proportional to Vphase2−Vo.  
         [0040]    Operation of the multi-phase synthetic ripple voltage regulator of the present invention may be readily understood with reference to the timing diagrams of FIG. 6. The uppermost portion of FIG. 6 shows a master ripple waveform  400 , which exhibits a sawtooth behavior with respect to the upper and lower thresholds Vupper and Vlower, respectively. The middle portion of FIG. 6 shows phase1 and phase2 ripple waveforms, which exhibit a sawtooth behavior with respect to the upper threshold Vupper. It is to be noted that the two instances of the Vupper threshold are in actuality at the same level. However, they have been separated in FIG. 6 in order to facilitate an illustration of the various ripple waveforms and, in particular, the times of occurrence of various events for those waveforms. This avoids a superimposed cluttering of the phase1 and phase 2 waveforms by the master ripple waveform. Finally, the lowermost portion of FIG. 6 shows a master clock (clk) signal that is produced at the QBAR output of flip-flop  230 , and the PWM1 and PWM2 waveforms produced at the Q outputs of output flip-flops  280  and  290 , respectively.  
         [0041]    Considering initially, the master ripple and the master clock waveforms, at time t0, the master ripple waveform is shown as decreasing and crossing the lower threshold Vlower. During the interval leading up to t0, the common terminal  241  of switch  240  is connected to input terminal  243 , so that a current proportional to ground (0V)−Vo, or simply −Vo is applied to capacitor  245 . Namely, the voltage V 245  across capacitor, which is the master ripple voltage, is decreasing during this interval. When (at time t0) this decreasing voltage crosses the lower threshold Vlower which is applied to the input  221  of comparator  220 , comparator  220  is tripped and resets flip-flop  230 . The latency between the actual crossing of the lower threshold Vlower and time t1 when flip-flop  230  resets (its QBAR output goes high) is due to second order circuit effects. When the QBAR output of flip-flop  230  goes high, the master clock (Master clk) goes high, and sequence logic  270  couples this output to the set input of the PWM1 output flip-flop  280 , so that its Q output  281  (which represents the PWM1 waveform) goes high at time t1.  
         [0042]    The change in state in the QBAR output of flip-flop  230  switches the connection of switch  240  to input  242 , so that the output of transconductance amplifier  250  is monitored by the hysteretic comparator circuitry. During a time interval beginning with t1, transconductance amplifier  250  produces an output current that is proportional to the difference between its inputs, namely proportional to Vin−Vo. This current is applied to capacitor  245 , so that as capacitor  245  is charged, its voltage (Master ripple) increases, as shown between time t1 and t2. Eventually, the increase in the master ripple voltage will exceed the upper threshold Vupper, causing comparator  210  to trip and set flip-flop  230 . It may be again noted that due to second order latency effects, the time t2 associated with the resetting of flip-flop  230  is slightly delayed relative to the actual instant at which the master ripple voltage crosses the upper threshold voltage Vupper.  
         [0043]    With flip-flop  230  now set, its QBAR output goes low at time t2, and remains there until it is again reset by comparator  220 , as described above. During the interval subsequent to time t2, with flip-flop  230  being set, switch  240  connects input  243  to its common terminal  241 , so that a negative current proportional to −Vo is again supplied to capacitor  245  by transconductance amplifier  260 , causing the master ripple voltage across capacitor  245  to decrease, as shown by the negative slope of the master ripple waveform. Eventually, at time t4, the master ripple waveform again crosses the lower threshold Vlower, so that comparator  220  is again tripped and resets flip-flop  230 . When the QBAR output of flip-flop  230  goes high, sequence logic  270  couples this output via output port  272  to the set input of the PWM2 output flip-flop  290 , so that its Q output  291  (the PWM2 waveform) goes high at time t4.  
         [0044]    The reset state of flip-flop  230  switches the connection of the common terminal  241  of switch  240  to its input  242 , so that the output of transconductance amplifier  250  is now monitored by the hysteretic comparator circuitry. During a new time interval beginning with time t4, transconductance amplifier  250  produces an output current that is proportional to the difference between its inputs, namely proportional to Vin−Vo. Again, as described above, this current is applied to capacitor  245 , so that capacitor  245  is charged causing its voltage Master ripple to increase, as shown in the interval between times t4 and t5. Eventually, this increase in Master ripple voltage will exceed the upper threshold Vupper, causing comparator  210  to trip, setting flip-flop  230 .  
         [0045]    With flip-flop  230  again set, its QBAR output goes low at time t5, and remains there until it is once again reset by comparator  220 , as described above. During the interval subsequent to time t5, with flip-flop  230  set, switch  240  reconnects input  243  to its common terminal  241 , so that a negative current is again supplied to capacitor  245  by the transconductance amplifier  260 , causing the master ripple voltage across capacitor  245  to decrease, as shown by the negative slope of the master ripple waveform during the time interval t5-t7. Eventually, at time t7, the master ripple waveform crosses the lower threshold Vlower, so that comparator  220  is again tripped and resets flip-flop  230 . When the QBAR output of flip-flop  230  again goes high, sequence logic  270  recouples this output via output port  271  back to the set input of the PWM1 output flip-flop  280 , so that its Q output  281  (and thereby the PWM1 waveform) goes high at time t7. This above process is repeated for subsequent cycles, as shown.  
         [0046]    Although the master ripple generator portion of the circuit directly controls the generation of the master clock and the rising edges of the PWM1 and PWM2 waveforms, its does not directly control the falling edges of the PWM1 and PWM2 waveforms. The falling edges are controlled by the phase1 and phase 2 ripple waveforms, as will described below. It should be noted, however, that the master ripple generator serves to control the frequency of the master clock and thereby the ripple voltages, since its generation is dependent upon the input and output voltages. Increasing the input voltage Vin increases the magnitude of the current (Vin−Vo) supplied by transconductance amplifier  250  to capacitor  245 , and thereby reduces the time required for the master ripple voltage across capacitor  245  to reach the upper threshold voltage Vupper. Conversely, decreasing the output voltage Vo not only increases the magnitude of the current (Vin−Vo) supplied by transconductance amplifier  250 , but increases the magnitude of the negative current supplied by transconductance amplifier  260 , the latter being effective to reduce the time required for the master ripple voltage across capacitor  245  to reach the lower threshold voltage Vlower.  
         [0047]    As pointed out above, transconductance amplifiers  320  and  330  produce output currents Phase1 ripple and Phase2 ripple that are respectively proportional to Vphase1−Vo and Vphase2−Vo, with the voltages Vphase1 and Vphase2 corresponding to the voltages at nodes  35  of the converter circuits associated with respective phases of the multiphase DC-DC converter. Considering first the Phase1 ripple waveform, the phase1 ripple waveform is shown as decreasing and the waveform continues to decrease until the master ripple voltage crosses the lower threshold, at time t0, so that comparator  220  is tripped and resets flip-flop  230 . As described above, due to second order latency effects, flip-flop  230  is reset at time t1, at which time sequence logic  270  drives the set input of the PWM1 output flip-flop  280 , so that its Q output  281  and thereby the PWM1 waveform goes high. With the PWM1 waveform going high, the Vphase1 voltage at node  35  of its associated DC-DC converter is driven high, so that transconductance amplifier  320  begins to charge capacitor  305  with a current proportional to Vphase1−Vo, whereby the voltage across capacitor  305  increases, as shown by the positive slope portion of the phase1 ripple voltage beginning at time t1. Eventually, this increasing phase1 ripple voltage, which is applied to the inverting (−) input  302  of comparator  300  crosses the upper threshold voltage Vupper, which is applied to the non-inverting input  301  of comparator  300 . When this happens, and taking into account second order latency effects, comparator  300  is tripped at time t3, and therefore drives the reset input of PWM1 output flip-flop  280 . With flip-flop  280  being reset by comparator  300  at time t3, the Q output  281  of flip-flop  280  is now driven low, causing the PWM1 waveform to go low. The PWM1 waveform will remain low until flip-flop  280  is again set at time t7 as described above. During the interval from t3 to t7, the relatively low phase1 voltage derived from phase node  35  causes transconductance amplifier  320  to apply a negative current (on the order of −Vo) to capacitor  305 , so that the phase1 ripple voltage waveform is continuously decreasing until the next cycle for PWM1.  
         [0048]    The Phase2 ripple waveform operates in the same manner as the Phase1 waveform, described above, except that it is every other master clock cycle relative to the Phase1 waveform. Namely, just prior to time t4, the phase2 ripple waveform is decreasing and the phase2 ripple waveform continues to decrease until the master ripple voltage crosses the lower threshold, so that comparator  220  is tripped and resets flip-flop  230 . As described above, due to second order latency effects, flip-flop  230  is reset at time t4, at which time sequence logic  270  drives the set input of the PWM2 output flip-flop  290 , so that its Q output  291  and thereby the PWM2 waveform goes high. With the PWM2 waveform going high, the Vphase2 voltage at node  35  of its associated DC-DC converter is driven high, so that transconductance amplifier  330  begins to charge capacitor  315  with a current proportional to Vphase2−Vo, which increases the voltage across capacitor  315 , as shown by the positive slope portion of the phase2 ripple voltage beginning at time t4. Eventually, this increasing phase2 ripple voltage, which is applied to the inverting (−) input  312  of comparator  310  crosses the upper threshold voltage Vupper, which is applied to the non-inverting input  311  of comparator  310 . When this happens, and taking into account second order latency effects, comparator  310  is tripped at time t5, and therefore drives the reset input of PWM2 output flip-flop  290 . With flip-flop  290  being reset by comparator  310  at time t5, the Q output  291  of flip-flop  290  is now driven low, causing the PWM2 waveform to go low. The PWM2 waveform will remain low until flip-flop  290  is eventually again set by the next alternating cycle of the master clock, subsequent to that occurring between t7 and t8. During the next interval beginning with time t6, the relatively low phase2 voltage derived from the phase node  35  causes transconductance amplifier  330  to apply a negative current (on the order of −Vo) to capacitor  315 , so that the phase2 ripple voltage waveform is continuously decreasing until the next cycle for PWM2.  
         [0049]    In accordance with a first alternative approach, the master ripple waveform produced across capacitor  245  may be created by a discharge and reset technique, using a single comparator as shown in FIG. 7, and the associated timing diagram of FIG. 8. At a time to, capacitor C 245  is discharged by a current proportional to Vo. When the voltage across capacitor C 245  drops below or crosses the threshold Vlower at t1, the output of comparator  80  and a one-shot  82 , shown as MSLCK, close the switch and reset the voltage across capacitor C 245  to the value of the upper voltage rail Vupper during the interval from t3 to t4. It should also be noted that a pair of master ripple capacitors may be employed in the place of the signal master capacitor C 245 . In this case the two capacitors alternately discharge from Vupper to Vlower, which serves to eliminate the reset interval (from t3 to t4).  
         [0050]    [0050]FIGS. 9 and 10 diagrammatically illustrate an alternative technique to produce the output signal PWM1. This same circuit may be applied to any of the other phases in a multiphase application. At time t0 in the timing diagram of FIG. 9, the signal CLK 1  ( 271 ) from the sequence logic causes the output port (PWM1) of flip-flop  280  to go high, and a switch  350  is turned on. The ripple capacitor voltage across capacitor C RIP  increases by a charge current that is proportional to (Vin−Vo). At time t1, the phase1 ripple voltage crosses the upper voltage threshold Vupper, and the comparator RRCMP resets flip-flop  280 , causing the PWM1 output to change state (go low). During the interval from t1-t2, the voltage across capacitor C RIP  decreases by a discharge current proportional to Vo.  
         [0051]    A beneficial feature of the present invention, particularly in connection with multiphase systems, is the fact that it varies the converter&#39;s switching frequency in response to load changes, something which the prior art does not do. In contrast, the prior art hysteretic converter of FIG. 1, described above, actually slows down the switching frequency during a load step (increase). This load step causes a depressed output voltage, which has the effect of turning on the high side or upper FET  30 , and leaves that FET on, until the output voltage at node  55  increases to the upper hysteretic set point, shown at  71  in FIG. 2. This means that such a control method is problematic in a multiphase system, where a single converter channel must pick up the full load current unit it can drive the output voltage above the upper hysteretic set point. As a consequence, a full load transient applied to a multiphase converter (such as a three-phase converter) results in one power channel having to deliver three times its steady state power.  
         [0052]    In accordance with the present invention, this problem is obviated by increasing the converter&#39;s switching frequency in response to a load step. This may be understood with reference to the block diagrams of FIGS. 3 and 5, described above, and the timing diagrams of FIGS. 11, 12 and  13 . In particular, for a load step (increase), the voltage at the output node  55  will initially decrease, which is fed back to input  131  of the error amplifier  130 . This decrease in the voltage at error amplifier  131  creates a larger differential across the error amplifier input and therefore a higher Vupper value produced at its output  133 . This transitional increase in the value of Vupper applied to input  211  of amplifier  210  in FIG. 5 (and that of its associated voltage value Vlower applied to the input  221  of amplifier  220 ) is shown in FIG. 11. As can be seen therein, the master ripple will now encounter the Vupper and Vlower references more frequently, so that the Q output of flip-flop  230  will produce a master clock more frequently, as shown in FIG. 12. FIG. 13 graphically illustrates the change in frequency between a first relatively steady state having a frequency on the order of 289 KHz, followed by a transition (during the transient state) to a frequency on the order of 560 KHz which, in turn, is followed by a further steady state frequency on the order of 300 KHz.  
         [0053]    It may be noted that the master clock signal initiates the PWM pulse which turns on the upper FET of the next successive power channel of the multiphase system, with the next power channel being selected by the sequence logic  270 . Increasing the switching frequency means each successive power channel will pick up the load sooner than it does during steady state, so that all of the power channels participate in picking up a power of the transient load current.  
         [0054]    An additional advantage of this method results for transient load steps that are less than full load. This may be contrasted with having to synchronize all of the power channels to turn-on the upper FET in each power channel in response to a load transient. With a less than full load transient, the resulting voltage is likely to overshoot the target regulation voltage. The present invention provides a relative smooth response to any magnitude transient.  
         [0055]    As will be appreciated from the foregoing description, by applying functionality of the transconductance amplifier and hysteretic comparator architecture disclosed in the above-referenced &#39;787 application to a multiphase DC-DC voltage generator, the present invention is able to realize a new and improved circuit arrangement for synchronizing a plurality of synthetic ripple voltage generators, that generate artificial or synthesized ripple voltage waveforms for controlling switching operations of a multiphase DC-DC voltage converter.  
         [0056]    While we have shown and described an embodiment in accordance with the present invention, it is to be understood that the same is not limited thereto but is susceptible to numerous changes and modifications as known to a person skilled in the art, and we therefore do not wish to be limited to the details shown and described herein, but intend to cover all such changes and modifications as are obvious to one of ordinary skill in the art.

Technology Classification (CPC): 7