Abstract:
A SAR converter having enhanced performance by virtue of effectively pre-loading the SAR&#39;s most significant bits with a value that makes the associated DAC output almost equal to the signal to be converted. A normal SAR conversion is then completed with the SAR bits that have not been pre-loaded. The value used to pre-load the most significant bits of the SAR is preferably obtained from a low-resolution, high-speed converter, such as a flash. The range of DAC bits used in the normal SAR part of the conversion may be increased such that errors up to a certain magnitude in the high-speed converter can be corrected. A method for performing an enhanced SAR conversion is also described.

Description:
FIELD OF THE INVENTION 
     This invention relates generally to analog-to-digital conversion and in particular to successive approximation A-to-D converters, and is more particularly directed toward a successive approximation A-to-D converter in which the most significant bits of the SAR are pre-loaded with a value nearly equal to the desired result. 
     BACKGROUND OF THE INVENTION 
     Successive approximation ADC is the name commonly given to an analog-to-digital conversion process in which digital approximations of the input analog voltage are determined on the basis of a binary search. A digital value stored in an n-bit successive approximation register (SAR) is input to a digital-to-analog converter, and a decision is made as to whether the value in the SAR represents an analog voltage that is higher or lower than the input analog value. 
     The initial value of the SAR is conventionally set to one-half the number that can be represented in the n bits of the SAR. To be more precise, an n-bit register can contain a value of 2 n −1, but for purposes of successive approximation, the initial value has the most significant bit set and the others cleared, which translates into a value of 2 n /2. If this comparison reveals that the digital approximation is indeed lower than the input voltage, the bit that was initially set remains set, the bit of next greater significance is also set, and another trial commences. If on the other hand, the SAR value is greater than the input analog voltage, the bit that was set for that trial is cleared, the bit of next greater significance is set, and another trial commences. It can be appreciated from this example why a successive approximation approach bears such a similarity to a binary search procedure. 
     Each bit of the SAR is set or cleared based upon a trial, so the conversion process requires only “n” trials to reach completion. SAR-type algorithms achieve conversion in much less time than a ramp-up technique. A ramp-up type of conversion, for example, requires that the input register of the D/A converter “count-up” by increments of 1 until the analog value is reached. Since each increment of the input register requires a trial to determine whether the analog input level has been reached, many trials may have to be performed before a successful conversion is achieved. There are other types of conversion systems that are even faster than SAR ADC, such as pipeline, flash, and half-flash, but these techniques require much more power than the SAR approach, and are thus unsuitable in many applications. 
     The main difficulty in AID conversion generally is settling time. The digital value that is written to the input register (the SAR register in a successive approximation system) produces an analog output at the D/A converter which must be allowed to settle completely before a comparison is performed in order to guarantee system accuracy. 
     As noted, a conventional SAR converter consists of only a DAC (digital-to-analog converter), a SAR register, control logic, and a single comparator. To generate an N-bit conversion result, each component of the converter is used (or updated) N times in a series of what are known as bit trials. Since only one comparator is used, the converter linearity is only limited by the accuracy of the DAC. The conversion time is dominated by the settling time of the DAC, which has to settle to system accuracy every bit trial. Since it is now possible to make very accurate DACs, SAR converters can be made to be extremely accurate. They also require very few analog components. However due to the serial way by which the result is derived, they are inherently slow. 
     Accordingly, a need arises for an ADC that occupies a relatively small amount of valuable integrated circuit real estate, has a high accuracy, and reaches a conversion result rapidly. 
     SUMMARY OF THE INVENTION 
     These needs and others are satisfied by the SAR ADC of the present invention. While successive approximation SARs are extremely efficient at moderate throughput rates of up to 1 MHz (megahertz), for example, the large number of clock cycles necessary to implement a single conversion makes it very difficult to implement significantly higher speed converters. Keeping in mind that a conventional SAR converter includes a DAC and an associated SAR register, a way to dramatically reduce the number of clock cycles required is to pre-load the SAR register&#39;s most significant bits with a value that makes the DAC output almost equal to the signal to be converted. A normal SAR conversion is then completed with the SAR bits that have not been pre-loaded. 
     The value used to pre-load the most significant bits of the SAR can be obtained from a low-resolution, high-speed converter, such as a flash. However, with a binary weighted SAR converter architecture, the accuracy of the high-speed converter must be at least the same as the required accuracy of the whole converter. But if the range of the DAC bits used in the normal SAR part of the conversion is increased, then errors up to a certain magnitude in the high-speed converter can be corrected. The range can be increased by adding what are sometimes known as redundant bits. These redundant bits are extra bits that would not strictly be required if no inaccuracy existed in the high-speed converter. 
     In accordance with one aspect of the present invention, an improved successive approximation analog-to-digital converter that produces an output digital representation of an input signal to be converted, includes a successive approximation register driving a first digital-to-analog converter, and a comparator to which both the output of the first digital-to-analog converter and the input signal are applied. The improvement comprises a second digital-to-analog converter driven by an analog-to-digital converter, wherein the output of the second digital-to-analog converter is summed with the output of the first digital-to-analog converter prior to being applied to the comparator. Preferably, the analog-to-digital converter driving the second digital-to-analog converter performs a coarse conversion of the input signal to be converted and propagates the coarse conversion result to the second digital-to-analog converter prior to the successive approximation conversion. 
     In one form of the invention, the analog-to-digital converter driving the second digital-to-analog converter may incorporate, at least in part, a parallel analog-to-digital converter. The parallel analog-to-digital converter may comprise, at least in part, a flash converter or a pipeline converter, for example. 
     In another form of the invention, the first digital-to-analog converter has an associated full-scale range derived by subtracting negative full-scale value from positive full-scale value, and the full-scale range is constrained to be greater than a value derived by subtracting the weight of an LSB of the first digital-to-analog converter from the weight of an LSB of the second digital-to-analog converter. 
     In still another form of the invention, the first digital-to-analog converter has a full-scale range at least equal to a value derived by subtracting the weight of an LSB of the first digital-to-analog converter from the weight of an LSB of the second digital-to-analog converter, and adding the magnitudes of the analog-to-digital converter&#39;s most positive code transition error plus the magnitude of the analog-to-digital converter&#39;s most negative code transition error, such that potential errors introduced by analog-to-digital converter code transition errors are eliminated from the output digital representation. Of course, either of the analog-to-digital converter&#39;s most positive or most negative code transition errors may be zero. 
     Preferably, negative code transition errors are effectively eliminated by shifting all code transitions in a positive direction by an amount at least equal to the magnitude of the largest negative code transition error. Negative code transition errors may be effectively eliminated by shifting comparator thresholds, for example, or by offsetting the analog input to the analog-to-digital converter in the negative direction by an amount at least equal to the magnitude of the largest negative code transition error. 
     In yet a further form of the invention, the analog-to-digital converter may be implemented, at least in part, using switch capacitor techniques, where the offset is achieved by means of at least one additional capacitor and at least one additional device that functions as a switch. In still another form, the digital output from the analog-to-digital converter may first be shifted by adding a fixed offset before being applied to the first digital-to-analog converter. 
     In still a further form of the invention, the full-scale range of the first digital-to-analog converter is extended by at least an amount equal to the sum of the magnitudes of the positive and negative code transition errors of the analog-to-digital converter driving the second digital-to-analog converter. Preferably, the full-scale range of the first digital-to-analog converter is extended in the direction of the negative full-scale value by application of one or more fixed offsets. One or more fixed offsets is applied prior to completion of a successive approximation conversion, regardless of comparator output. 
     It is preferred that the total value of the fixed offset (or offsets) be at least as negative as the analog-to-digital converter&#39;s most negative code transition error. The fixed offset or offsets may be provided by a third digital-to-analog converter. 
     The first digital-to-analog converter may be binary weighted or non-binary weighted. In either case, it is preferred that the MSB of the first digital-to-analog converter have a weight that is at least equal to the weight of the LSB of the second digital-to-analog converter. 
     In one form of the invention, the output digital representation is derived from the digital input to the first digital-to-analog converter and the digital input to the second digital-to-analog converter, by providing the digital inputs to the first and second digital-to-analog converters, and providing a digital representation of the total value of all fixed offsets, as three inputs to recombination logic that provides the output digital representation. Preferably, the recombination logic operates on the three inputs to provide the output digital representation that corresponds to the sum of the predetermined bit weights in the first and second digital-to-analog converters, plus the total weight of the fixed offset (or offsets). 
     In accordance with another aspect of the present invention, a successive approximation analog-to-digital converter that produces an output digital representation of an input signal to be converted comprises a successive approximation register driving a first digital-to-analog converter, a comparator to which both the output of the first digital-to-analog converter and the input signal are applied, and a second digital-to-analog converter driven by an analog-to-digital converter that performs a coarse conversion of the input signal to be converted, and propagates the coarse conversion result to the second digital-to-analog converter prior to the successive approximation conversion. The output of the second digital-to-analog converter is summed with the output of the first digital-to-analog converter prior to being applied to the comparator. 
     In accordance with yet another aspect of the present invention, a method for performing an analog-to-digital conversion to obtain a digital representation of an input signal comprising the steps of performing a first conversion of the input signal to obtain an estimate of at least the most significant portion of the digital representation, the first conversion being performed, at least in part, utilizing a parallel analog-to-digital converter, deriving an initializing value for a successive approximation register of a successive approximation analog-to-digital converter from the estimate, and performing a second conversion of the input signal, using the successive approximation analog-to-digital converter, to determine the remaining bits of the digital representation. 
     Further objects, features, and advantages of the present invention will become apparent from the following description and drawings. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 illustrates a successive approximation analog-to-digital converter of the prior art; 
     FIG. 2 depicts a type of parallel converter known in the art; 
     FIG. 3 illustrates an ADC architecture in which a SAR and a high-speed converter are combined; 
     FIG. 4 shows the ideal characteristics for the 4-bit converter of FIG. 3, where there are no errors in the flash converter; 
     FIG. 5 illustrates the characteristic for the converter of FIG. 3 where ADC 1  converter errors occur; 
     FIG. 6 depicts an ADC system in accordance with the present invention; 
     FIG. 7 is a timing diagram for the converter of FIG. 3; 
     FIG. 8 is a timing diagram for the converter of FIG. 6; and 
     FIG. 9 is an additional illustrative timing diagram for the converter of FIG.  6 . 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     There is described herein a SAR ADC with pre-loaded SAR registers that offers distinct advantages when compared to the prior art. A conventional successive approximation analog-to-digital converter of the prior art is illustrated in FIG.  1  and generally depicted by the numeral  100 . A successive approximation register  105 , which is “n” bits wide, and loaded by control logic  106 , is coupled to the input register of an n-bit digital-to-analog converter (DAC)  104 . The DAC output, in turn, is coupled to one of the inputs of a comparator  102 . The analog input voltage to be measured, V IN    101 , is coupled through a switch  109  to a sample and hold amplifier  107 . A switch  109  is interposed between the analog input signal  101  and the input of the amplifier  107 . A track/hold control signal from the control logic  106  controls the state of the switch  109 . 
     When the switch  109  is ON, the input signal  101  is allowed to reach the input of the amplifier  107 , with a capacitor  108  to ground. In this mode, the capacitor  108  is allowed to charge to the level of the input signal  101 . Thus, this configuration represents the TRACK mode of operation of the circuit. 
     When the switch  109  is OFF, on the other hand, the input signal path to the amplifier  107  is interrupted, and the voltage on the holding capacitor  108  is applied to the amplifier. This condition corresponds to the HOLD mode of operation. The logic state of the output  103  of the comparator  102  indicates the relative amplitude of the DAC output voltage with respect to the input analog voltage. 
     A conventional SAR A/D  100  “finds” the value of the analog input signal  101  using a technique that is very similar to a binary search. Initially, the comparison value provided to the SAR  105  has only the most significant bit set and the remaining bits cleared. The analog input voltage  101  is either less than or greater than the comparison value derived from the SAR contents, in this initial case half of the voltage range. For each subsequent trial, less significant bits in the SAR are set and the comparison trials are repeated, but the “windows” of comparison always correspond to the voltage range between zero volts and the comparison level (a binary weighted fraction of the full scale voltage), and between the comparison level and the full scale voltage. 
     One of the disadvantages of the binary weighted voltage search algorithm associated with conventional successive approximation techniques is that, if an error happened to occur on a given trial, there is no provision for error correction. In other words, a bit set or cleared in error in the SAR always remains set or cleared, and that error persists through to the end of the conversion. 
     FIG. 2 depicts a type of parallel converter commonly known as a “flash” converter, in this example having a Gray code output, and generally depicted by the numeral  200 . As is well-known, in a Gray code system only one output bit changes state at any one time. This has the advantage of minimizing any noise spikes that might propagate through the system due to multiple output stage switching. 
     The flash converter  200  includes 2 n −1 comparators  201 , biased 1 LSB apart, starting with +½ LSB. Each comparator  201  has a unique threshold voltage that is established through a resistor-string voltage divider  205  supplied by a reference voltage V REF    204 . For an input voltage V IN    202  of zero volts, all comparators are OFF. As the input voltage  202  increases, an increasing number of comparators  201  will change state. The comparator outputs are then coupled to logic gates  203  at the converter output, in order to generate the desired output coding (Gray code in the example). Of course, binary outputs, or another desired coding scheme, could be easily implemented. 
     The principal advantage of the flash converter is speed. The speed of the system is directly attributable to the fact that conversion occurs in parallel, and is limited only by the speed of the comparators and output logic. This type of parallel conversion is the fastest conversion approach. Of course, the parallel conversion is a mixed blessing. If higher resolution is desired, the number of elements required increases rapidly, until a point is reached where the converter uses more power and occupies more real estate than a particular design can tolerate. 
     As discussed above, SAR converters tend to be slow due to the serial fashion in which conversion results are derived. Flash converters, as noted, represent the opposite extreme in terms of converter architectures. An N-bit flash converter consists of 2 N −1 comparators which, in order to generate a conversion result, are all operated simultaneously. The accuracy of the converter is determined by the accuracy of the individual comparator thresholds. Since the speed of the comparator is much faster than the settling time of the DAC, and because all comparators are operated simultaneously, flash converters are much faster than SAR converters. However, they are expensive (in terms of analog components) and, because it is difficult to make high accuracy comparators, they tend not to be very accurate. 
     The large number of comparators required to make a flash converter, as well as their accuracy limitations, tend to restrict their use to low- to medium-resolution applications. However, since a comparator cell can be quite compact, up to about 6-bit resolution converters can be made in what, for many applications, is an acceptable die area. 
     FIG. 3 illustrates an ADC architecture in which a SAR converter  301 ,  304 ,  306  and a high-speed converter (ADC 1 )  302  are combined. The flash converter  302  generates the high order bits, which control DAC 1   303 , and the SAR converter, comprising the SAR register  301  and DAC 2   304 , generates the low order bits. DAC 1   303  and DAC 2   304  are combined in a summer  305  at the input to a comparator  306 . DAC 1   303  can be thought of as the most significant bits of the SAR converter, which, instead of being controlled by the SAR register, are in this new architecture controlled by the flash converter. 
     As an example, consider a 12-bit converter in which 6 bits are generated by the parallel converter ADC 1   302 , and 6 bits are generated by the SAR converter  301 ,  304 . Since the conversion time is dominated by the speed of the SAR converter  301 ,  304 , the overall conversion time is almost half that of a conventional 12-bit SAR converter. This architecture has one significant drawback. The accuracy of the conversion is limited by the accuracy of ADC 1   302 . If the accuracy of ADC 1   302  is only ±½ LSB ADC1 , then the accuracy of the final conversion result will be only ±32 LSB. This, of course, is unacceptable. 
     The positive and negative full-scale values for the SAR converter  301 ,  304 ,  306  of FIG. 3 are 2 n −1 and 0. For the SAR converter to be able to correct for the errors in ADC 1  the positive full-scale of the SAR must be increased to a value=(2 n −1+E pos ), and the negative full-scale must be decreased to a value=E neg , where E pos  is ADC 1 &#39;s most positive code transition error, and E neg  is ADC 1 &#39;s most negative code transition error. 
     In order to appreciate more fully why the additional range is required, one must look at the output voltages from DAC 1  and DAC 2 , at the end of the conversion, as a function of the input voltage  307  to the comparator  306 . FIG. 4 shows the ideal characteristics for a 4-bit converter where there are no errors in the flash converter. Both the flash and SAR converters are 2-bit in this example. Stair-step waveform  401  (bold solid line) represents the output of the flash converter DAC 1   303 , while waveform  402  (dashed line) represents the combined outputs of DAC 1  and DAC 2  at the output  308  of the summer  305 . 
     It should be noted that V dac −V input  is always in the range from 0 to −1 LSB. The reason why this value is never greater than 0 is that, while the SAR part of the converter is active, whenever a set bit results in V dac −V input  being greater than zero, that bit is then rejected. It is important that the difference value at the comparator input is always in the range 0 to −1 LSB, as any value outside this range corresponds to a conversion error. This is because there is a direct relationship between the digital value at the input to the DAC (which is the conversion result at the end of the conversion) and the DAC outputs. 
     FIG. 5 illustrates the same characteristic for a converter where the ADC 1  converter&#39;s 1 st  and 2 nd  transitions are in error by −E 1  and +E 2  respectively. The DAC 1  output is represented by waveform  501  (bold solid line), while the combined DAC 1 +DAC 2  output corresponds to waveform  502  (dashed line). The ideal DAC 1  output is represented by dashed line  503 . For voltages just above an input voltage V a , V dac −V input =E 1 . To bring V dac −V input  to within the required range of 0 to −1 LSB, DAC 2  would need to be able to generate a value of −E 1 . 
     Similarly, for an input voltage just below V b , V dac −V input =−1 LSB −E 2 . To bring V dac −V input  to within the range from 0 to −1 LSB, DAC 2  would need to be able to generate an additional voltage above DAC 2 &#39;s normal full scale value of +E 2 . 
     FIG. 6 depicts an ADC system in accordance with the present invention where the positive and negative full scales of the SAR converter  605 ,  606  have been extended to allow correction of errors in ADC 1   302 . The negative fall scale of the SAR has been reduced to −2 n  by the inclusion of the fixed offset DAC  601  and the positive full scale value has been increased to (2 n −1+2 n ) by the addition of a higher order bit B n+1  in DAC 2   605 . 
     The output of the fixed offset DAC  601  is combined with the output of DAC 2   605  in a first summer  602 . This combined result is then added to the output of DAC 1   303  in second summer  603 . The additional range provided through the inclusion of the fixed offset and the additional bit in DAC 2   304  allows E pos  and E neg  to be as large as +2 n  and −2 n  respectively, but still to be corrected during the SAR part of the cycle. Since the ranges of the 2 DACs now overlap, an additional digital block  604 , including recombination logic, is required to combine the digital inputs of the DACs and to account for the offset introduced by the fixed offset DAC  601 . 
     The digital block  604  requires only adders  607 ,  608  to perform the combination of the 2 DAC inputs and the offset value  609 . Using modern processes, these digital logic circuits can be made to occupy a die area significantly smaller than that necessary for analog circuits. The apparent multiplication  610  by 2 n+1  is achieved simply by shifting bus A  611  so that the LSB of bus A is applied to adder input 2 n+1 . Modifications to the logic are possible in which the digital recombination is only done at the end of the conversion, or, alternatively, some of the recombination can be performed during the conversion. Also, the order of addition of the three digital values may be changed. 
     It should be noted here that the fixed offset (or offsets) that extends the negative full-scale of DAC 2  ( 605  in FIG. 6) in the negative direction has to be at least as negative as the analog-to-digital converter&#39;s  302  most negative code transition error to avoid any consequent conversion error. However, if the result, generated by the analog-to-digital converter  302  has only positive code transition errors, then no negative fixed offset or offsets are required. This can be arranged in a number of ways: 
     First, the ADC  302 , as so far described, would typically be expected to have both positive and negative code transition errors. By shifting all the code transitions up by an amount at least equal to the magnitude of the largest negative code transition error, the negative code transition errors would be eliminated. For a flash converter, the shifting of the code transitions can be achieved by simply shifting the comparator thresholds. 
     As a second possibility, the analog input  307  to the ADC could be offset in the negative direction by an amount at least equal to the magnitude of the largest negative code transition error. Switch capacitor techniques allow this to be achieved by means of additional capacitor(s) and switch(es). 
     In yet another alternative, digital logic can be inserted between the unmodified ADC&#39;s  302  output and DAC 1 &#39;s  303  input. The digital logic would generate a digital value that is shifted down by an amount at least equal to the magnitude of the largest negative code transition error. 
     Each of the above techniques also has the effect of increasing the positive code transition error of the overall ADC. However, eliminating the negative fixed offsets may result in a simplification of the overall architecture. 
     FIG. 7 is a timing diagram for the converter of FIG. 3, where both DAC 1   303  and DAC 2   304  are 2-bit converters (N=3 and n=1). The analog input voltage is represented by waveform  307  (alternating long and short dashes), while the DAC 1  output voltage is shown in waveform  701  (bold solid line). The dashed-line waveform  702  represents the combined DAC 1  and DAC 2  outputs. 
     On the rising edge of ADC 1  STROBE  703 , ADC 1  ( 302  in FIG. 6) converts the analog input and the result appears on DAC 1  input A 3,2    704 . Bit A 1  is then set in the SAR register (as shown in waveform  706 ), and, once the DAC has settled to sufficient accuracy, the comparator is strobed by control line COMP 1  STROBE  705 . Since the combined DAC output is larger than the analog input  307 , bit A 1  is reset. Bit A 0  is then set and the comparator is once again strobed, resulting in bit A 0  also being reset. It should be emphasized that the combined DAC output should be less than the analog input at the end of the conversion. Of course, during the bit trials, the combined DAC output may be larger than the analog input, but when this happens the trial bit is then rejected. 
     FIG. 7 shows that ADC 1 &#39;s first and second code transitions are in error by −2 and +2 LSB, and V input  is just above ADC 1 &#39;s first transition. The first transition error results in ADC 1  incorrectly generating an output 01 (rather than 00). Despite both bits of the SAR converter being subsequently rejected, the error cannot be recovered and the resulting conversion is in error by +2 LSB. The error is 2 LSB because V dac −V input  is 2 LSB, while it should be in the range 0 to −1 LSB. It should be noted from an inspection of bits A 3,2,1,0  (waveform  707 ) that the final result in FIG. 7 is 0100, while it should actually be 0010. 
     FIG. 8 is a timing diagram for the converter of FIG. 6, where DAC 1   303  is 2-bit and DAC 2   605  is 3-bit (N=1 and n=1). Waveform  307  (alternating long and short dashes) is once again the analog input voltage. Waveform  801  (bold solid line) represents the output of DAC 1  combined with the offset, while waveform  802  (short dashes) is the combination of DAC 1 , DAC 2 , and the offset voltage. ADC 1 &#39;s transition errors are the same as in FIG.  7 . Also, just as in FIG. 7, V input    307  is just above ADC 1 &#39;s first transition, represented by a dotted line  803 . 
     As shown, by rejecting all 3 bits in the SAR, ADC 1 &#39;s error can be recovered and a correct conversion results. FIG. 9 illustrates the timing of the converter of FIG. 6 where V input    307  (alternating long and short dashes) is just less than the second transition error represented by dotted line  903 . In the example of FIG. 9, all bits in the SAR converter are kept and again a correct conversion results. The waveform  307  is the analog input voltage in FIG. 9, as noted, while waveform  901  (the bold solid line) corresponds to the DAC 1  output plus the offset voltage, and waveform  902  (short dashed line) is the combined DAC 1  and DAC 2  outputs with the offset also applied. 
     There has been described herein a SAR ADC with pre-loaded SAR registers that offers distinct advantages when compared with the prior art. It will be apparent to those skilled in the art that modifications may be made without departing from the spirit and scope of the invention. For example, the SAR DAC described in conjunction with the present invention need not be binary weighted. Although binary weighted DACs are still the most common form of DAC used within SAR converters, non-binary weighted DACs are beginning to become more popular, particularly for high-speed SAR converters. 
     The reason for this is that non-binary weighted DACs are more error tolerant: whereas non-binary weighted DACs cannot recover from any bit being incorrectly kept or rejected, non-binary weighted DACs have a certain tolerance to such errors. For example, if a bit with a weight of 4 LSBs in a binary weighted DAC is wrongly rejected (due to insufficient settling time in the DAC, for instance), the DAC output at the end of the conversion may end up being as much as 5 LSBs less than the analog input. This difference should be in the range 0 to 1 LSB. By adding an additional “redundant” bit, also with a weight of 4 LSBs, even if the first bit is incorrectly reset to zero, then as long as the second 4-LSB bit is correctly kept, the error is completely recovered. 
     Since the DAC is no longer binary weighted, additional recombination logic is required to generate a true binary result. However, since the use of extra redundant bits allows lower performance analog circuits to be used, this trade-off in some cases is well worthwhile. In practice, to allow correction of both wrongly kept and rejected bits, one or more negative fixed offsets have to be switched in during the conversion. These offsets must be accounted for in the recombination logic. 
     The additional redundant bit(s) also have the effect of increasing the full-scale range of the DAC. Some or all of this additional range may be allocated to correction of bit decision errors in the successive approximation converter itself. To avoid overall conversion errors, the fill-scale range of the SAR converter DAC must be sufficient to allow correction of both code transition errors in the ADC and bit trial errors in the SAR converter itself. 
     It is also possible to implement digital-to-analog converters in a number of ways, specifically including switch capacitor techniques. It is even possible to construct a capacitor array that functions as both the sampling capacitor and the DAC. Such a system is described in detail in U.S. Pat. No. 4,129,863, issued Dec. 12, 1978 to Paul R. Gray et al., and fully incorporated by reference thereto as though fully set forth herein. In view of the modifications suggested above, and others that will occur to the reader skilled in the art, it is not intended that the invention be limited except as may be necessary in view of the appended claims.