Abstract:
A circuit for reducing power loss for a soft switching full bridge converter at light loads and enabling very high frequency operation without using a cold plate approach. The circuit preferably includes a resonant inductor and blocking inductor on the converter&#39;s primary side arranged so as to provide reduced losses for a zero voltage switching bridge converter. The circuit provides these benefits even for converters having a power transformer with very low leakage inductance. The circuit is not dependent on the presence of a high leakage inductance for the power transformer. The circuit can also be used in soft switching half bridge converters. The circuit can also be used in a hard switching full bridge or half bridge converter for achieving zero voltage switching at reduced cost with reduced losses at light load, if the duty cycle of the converter is set near fifty percent.

Description:
CROSS REFERENCE TO RELATED APPLICATION 
   This application claims the benefit of U.S. Provisional Application No. 60/493,632, filed Aug. 9, 2003, which is incorporated by reference herein. 

   FIELD OF INVENTION 
   The present invention relates in general to power converter circuits and more particularly to zero voltage switching (ZVS) half bridge and full bridge converters and a circuit that reduces the high power loss at very light loads for such converters operating at high frequency. 
   BACKGROUND OF INVENTION 
   The modern technology trend is towards higher density and lower profile electrical devices. This trend has driven a demand for improving the power density of power supplies. As a result, various techniques have been developed to increase the switching frequencies of the power supply in order to reduce the size and bulk of magnetic components and filtering elements. Problems experienced with increasing the operating frequencies include higher switching losses and the generation of worse electromagnetic interference (EMI). Since the switching losses in power semiconductors are directly proportional to the operating frequency, thermal management is also a big challenge, since the space saved by using smaller filtering components is more than offset by the need for larger heat sinks. 
   The development of a method of soft switching power supplies addressed most of the above problems. For the DC/DC conversion stage, the soft, zero voltage switching (ZVS) method for the power switches eliminated turn on losses. At the same time, this method improved EMI performance by lowering fast rising switching currents. Thus, the method significantly improved the efficiency of the power converter and enabled switching at higher frequencies. The demand for higher power density, however, is increasing unabated. New problems are surfacing as the soft, ZVS converter is being switched at ever higher frequencies. Conventional soft switching converters switching at high frequencies are exhibiting high power losses during light load conditions. As a result, although such converters are very efficient at full load, they are prone to failure at light loads. Many semiconductor manufacturers attribute this failure to various semiconductor phenomena, such as the reverse recovery speed of the MOSFET body diode, and the construction of the MOSFET&#39;s channel, etc. One other significant problem, however, is that it is difficult to charge and discharge the output capacitance of the MOSFET bridge switching elements at light loads when the converter is in the hard switching mode. This is because the energy stored in the resonant choke in such converters is very low and therefore cannot charge/discharge the output capacitance of the bridge switching elements. This is true for converters having a primary side resonant choke or a saturable choke on the secondary side. This problem exists in all known ZVS control techniques. In fact, this drawback at light load for conventional soft switching converters is even worse than for conventional hard switching bridge converters. 
     FIG. 1  illustrates a worst case scenario when the power supply is at no-load. At no-load, the only energy available for ZVS switching is the magnetizing current of the power transformer. Since this current is typically very low, it cannot charge/discharge the MOSFET switch capacitance in the required delay time. 
   As shown in  FIG. 1 , the circuit on the left side shows a bridge circuit  10  with a transformer primary winding. Since the transformer magnetizing current is very low at a no-load condition, negligible energy is stored in the primary winding or in any series resonant choke. The series resonant choke for the circuit in  FIG. 1  could be the leakage inductance of the transformer or any external inductance. Since the energy stored in the magnetizing inductance is very low, it cannot enable ZVS of the bridge elements, and thus, can be neglected in the analysis. The bridge circuit  20  on the right side in  FIG. 1  shows this worst case scenario where the transformer winding has no effect. 
   To analyze the circuit in  FIG. 1 , it is first assumed that an active diagonal is operating, e.g., switches Qa and Qd are on. At the end of the active period, switch Qd will turn off. The voltage across it will not rise since there is no charging current available. As a result, the voltage across Qd will remain close to zero during the dead time at the end of the active period. Thus, the MOSFET switch Qd output capacitance, shown as capacitor Cd, is at zero volts and fully discharged and the capacitance across switch Qc, shown as capacitor Cc, is fully charged at the Bulk+voltage shown. In typically operating soft switching converters, switch Qc will be turned on after a short delay. When switch Qc turns on, the energy stored in capacitor Cc is fully discharged in switch Qc. At the same time, as the lower end of the switch Qc rises to the Bulk+voltage level, the capacitance Cd of the lower switch Qd also gets charged through switch Qc. Thus, there are two kinds of resistive losses in switch Qc: one due to the discharging of capacitor Cc and other due to the charging of capacitor Cd. These losses result in power dissipation which is proportional to the operating frequency as represented in the following formula:
 
 P turn-on=(0.5 ×Cc× ( V bulk) 2   ×Fsw )+(0.5 ×Cd× ( V bulk) 2   ×Fsw )
 
   Where Fsw is the switching frequency. Assuming Ca=Cb=Cc=Cd:
 
 P turn-on= Cc ×( V bulk) 2   ×Fsw 
 
   These resistive losses, and the resulting power dissipation, may be tolerable at lower switching frequencies in the range of 100 kHz to 200 kHz. At much higher frequencies, e.g., above 400 kHz, however, these losses predominant such that the total power lost in the bridge switches at light loads exceeds the losses at full-load. This predominance is illustrated in  FIG. 1A  for a typical soft switching full bridge converter.  FIG. 1A  shows total losses in the entire converter versus the load percentage. At light loads, most of this total loss is due to losses in the bridge switching devices. 
     FIG. 2  is a circuit diagram of an exemplary prior art full bridge power converter  30  where a primary side resonant inductor is used for achieving soft, zero voltage switching. As is seen, a resonant inductor Lr is inserted in series with the primary of the power transformer. Inductor Lr could also be the parasitic leakage inductance of the transformer.  FIG. 2A  is a set of voltage and current waveforms illustrating the operation of the power converter in  FIG. 2 . A simplified representation of switches Qa, Qb, Qc, and Qd is shown in  FIGS. 2–4  such that the switch capacitances of the corresponding switches are not shown. The existence of the switch capacitances is well known in the art. For reference, the switch capacitances are as shown in bridge  20  for switches Qa, Qb, Qc, and Qd in  FIG. 1 . 
   During the active period of the switching diagonal, e.g., Qa and Qd are on, energy is stored in inductor Lr due to the primary current flowing through it. When one of the diagonal MOSFET switches (e.g., Qd) turns off, the energy stored in inductor Lr is used to charge that MOSFET&#39;s output capacitance and to discharge the output capacitance of the other MOSFET in the same vertical leg. As a result, ZVS action is achieved. 
   In addition to the fact that the circuit topology in  FIG. 2  has the drawback of losses at light load at higher frequencies, since the capacitance of each MOSFET switch is intrinsic and does not change with frequency, the size of the resonant inductor Lr is independent of frequency. As a result it may be quite large for a high frequency power supply. Inductor Lr is also lossy since it handles very high primary full load currents and its flux swings in both directions, generating high core losses. The series inductor in  FIG. 2  also introduces a delay, e.g., 200 nS, which reduces the available maximum duty cycle of the converter. This delay is a serious drawback at higher frequencies. 
     FIG. 3  is an exemplary prior art full bridge converter  40  where two saturable inductors, Ls 1  and Ls 2 , are connected in series with the secondary side&#39;s rectifier diodes. In operation, for an active transformer period when diagonal Qa–Qd is conducting, the dotted end of the secondary is positive and D 1  is forward biased, providing current to the output load through inductors Ls 1  and Lout. This current saturates inductor Ls 1 . At the end of the active period, Qd turns off and the secondary voltage starts to fall. Since Ls 1  is saturated and Ls 2  is in blocking mode since D 2  is reverse biased, this forces the current in output inductor Lout to keep flowing through the upper half of the secondary, i.e., D 1 -Ls 1 . This DC inductor current also has an AC component in the form of ripple current. The transformer action causes this ripple current to be reflected back to the primary side, which forces the primary current to keep flowing while achieving ZVS action. Similar ZVS action is repeated by Ls 2  in the next active period. 
   Although the circuit in  FIG. 3  achieves ZVS action satisfactorily at higher loads, it still has the drawback of losses at light load at higher frequencies. At frequencies above 200 kHz, for example, the core losses in the secondary side saturable cores of Ls 1  and Ls 2  are very high and could result in thermal runaway for the square loop amorphous cores typically used. The blocking effect of these saturable inductors also reduces the available duty cycle. 
     FIG. 4  shows a prior art full bridge converter  50  including two external resonant inductors Lr 1 , Lr 2  and two split capacitors C 1 , C 2  to generate a split bulk+voltage rail.  FIG. 4A  is a set of voltage and current waveforms illustrating the operation of the power converter in  FIG. 4 . In operation, when the diagonal full bridge devices, e.g., Qa, Qd, are in conduction, current flows in the respective inductor (Lr 1 , Lr 2 ) and energy is stored. At the end of the active period when the switch, e.g., Qd, turns off, the energy stored in the inductor is utilized to achieve the ZVS transition. 
   The prior art converter  50  shown in  FIG. 4  may provide zero voltage switching down to very light loads for all four full bridge MOSFETs, Qa, Qb, Qc and Qd, if the power transformer is non-ideal, i.e., has high leakage inductance, and thus may be able to address the problem of losses at light load. However, this circuit has several drawbacks. The circuit in  FIG. 4  requires the inclusion of two inductors, Lr 1 , Lr 2 , and two capacitors, C 1 , C 2 . The ripple current stress on the capacitors can be significant, such that capacitors of higher cost are required. Instead of using such costly capacitors, each of these bulk capacitors can alternatively be split into a series combination of two. The drawback of this solution is that this greater number of capacitors will occupy a larger volume, thereby creating an inefficient use of the available space. Another drawback of the circuit in  FIG. 4  is that any inequality between the values of C 1  and C 2  or between the values of Lr 1  and Lr 2  can create problems with the current mode control of the circuit. 
   Another drawback of this circuit is as follows. The circuit in  FIG. 4  can provide satisfactory ZVS transition from the active to the passive state. During the transition from the passive to the active state, however, the energy in inductors Lr 1  and Lr 2  can flow through the transformer and be transferred to the load instead of achieving ZVS transition of the passive to active leg. This drawback is lessened in applications having a large transformer leakage inductance, but for transformers with very low leakage inductance, this problem in the converter  50  shown in  FIG. 4  may result in some hard switching of one leg of the bridge. 
   The heat sink for most power supplies is designed to accommodate heat dissipation at full-load. Although a cooling fan is typically provided for the power supply, the fan is typically controlled such that fan speed is a function of the load. Thus, at light loads, the dissipation in bridge switches is higher than at full-load and much less cooling air is available. As a result, these devices may fail due to thermal runaway. Prior art devices have addressed this failure mode through a “cold plate” approach. In this approach, the bridge switches are mounted on the same large heat sink used for cooling the boost converter or secondary rectifiers. Since the power losses in the boost converter or output rectifiers are negligible at light loads, the large heat sink can handle the extra losses in the bridge devices at light load, and thereby avoid thermal runaway. This cold plate approach is inefficient and cannot meet more demanding efficiency requirements at light load conditions. The cold plate approach also complicates the construction of the power supply as several safety requirements must be met as well, e.g., requiring insulation on the secondary side, thus rendering this approach inconsistent with high density requirements. 
   A circuit is therefore needed which solves the above described drawbacks of losses at light load in high frequency soft switching power converters. 
   SUMMARY OF THE INVENTION 
   The present invention overcomes the drawbacks of power loss at light loads for soft ZVS half bridge and full bridge converters by providing a circuit that reduces the internal power losses of the soft switching full bridge converter at light loads and enables very high frequency operation without using a full cold plate approach. 
   An advantage of the present invention is that is provides design flexibility for practical applications. 
   Another advantage of the present invention is that it reduces the component cost since the ZVS inductor can be made using cheaper materials and dissipates lower power. 
   Another advantage of the present invention is that it lowers EMI at all load conditions. 
   Still another advantage of the present invention is that, unlike conventional ZVS converters, the value of the ZVS inductance reduces with increased operating frequency, thereby enabling higher density packaging by reducing component size. 
   Broadly stated, the present invention provides, in a DC-DC converter for providing substantially zero voltage switching (ZVS) having first and second input terminals to which an input DC voltage is coupled and two output terminals where the output DC power is provided, a bridge having a first and second switching leg, each leg comprising two controlled switches connected in series, each switch having a switch capacitance and a control input, each switching leg connected between the input terminals and having a junction point between its series-connected switches, a transformer having a primary winding and a secondary winding each having a first and second end, and a rectifier and output filter circuit coupled between the secondary winding and the output terminals, a circuit for reducing power losses at light loads and enabling very high frequency operation comprising a first inductor connected between the junction points for charging and discharging the switch capacitances; and a second inductor for providing high impedance to a sudden reversal of current having an end connected to the junction point of the first switching leg and another end connected to the first end of the primary winding, the second end of said primary winding is connected to the junction point of the second switching leg. 
   Broadly stated, the present invention also provides a DC-DC converter for providing substantially zero voltage switching (ZVS) having first and second input terminals to which an input DC voltage is coupled and two output terminals where the output DC power is provided comprising a bridge having a first and second switching leg, each leg comprising two controlled switches connected in series, each switch having a switch capacitance, each switching leg connected between the input terminals and having a junction point between its series-connected switches; a transformer having a primary winding and a secondary winding each having a first and second end; a rectifier and output filter circuit coupled between the secondary winding and the output terminals; a first and a second diode; a first inductor connected in series with the first diode between the junction points; and a second inductor connected in series with the second diode between the junction points; wherein the first diode has a cathode connected to an end of the first inductor and an anode connected to the junction point of the second switching leg and the second diode has a cathode connected to the anode of the first diode and an anode connected to an end of the second inductor; and a third inductor for providing high impedance to a sudden reversal of current having an end connected to the junction point of the first switching leg and another end connected to the first end of the primary winding, the second end of the primary winding is connected to the junction point of the second switching leg. 
   Broadly stated, the present invention also provides a DC-DC converter for providing substantially zero voltage switching (ZVS) having first and second input terminals to which an input DC voltage is coupled and two output terminals where the output DC power is provided comprising a bridge having a switching leg comprising two controlled switches connected in series, each switch having a switch capacitance, the switching leg connected between the input terminals and having a junction point between its series-connected switches, a transformer having a primary winding and a secondary winding each having a first and second end, a rectifier and output filter circuit coupled between the secondary winding and the output terminals; a capacitive voltage divider formed by a first and a second capacitor and connected between the input terminals, a first inductor connected between the junction point of the switching leg and a junction between the first and second capacitor; and a second inductor for providing high impedance to a sudden reversal of current having an end connected to the junction between the capacitors and another end connected to the first end of the primary winding, the second end of the primary winding is connected to the junction point of the switching leg. 
   Broadly stated, the present invention also provides a method of operating a bridge DC-DC converter for substantially zero voltage switching, the converter having first and second input terminals to which an input DC voltage is coupled and two output terminals where the output DC power is provided and which comprises a bridge having two switching legs, each leg comprising two controlled switches connected in series, each switch having a switch capacitance and a control input, each switching leg connected between the input terminals and having a junction point between its series-connected switches, a transformer having a primary winding and a secondary winding each having a first and second end, and a secondary circuit for deriving an output of the converter from the secondary winding, comprising the steps of supplying substantially complementary control signals to the control inputs of the switches in the first switching leg so that the corresponding switches conduct alternately with dead times therebetween; supplying relatively phase shifted substantially complementary control signals to the control inputs of the switches in the second switching leg so that the corresponding switches conduct alternately with dead times therebetween; and during the dead time when one of the switches in the first switching leg has been turned off, charging the switch capacitance of the turned-off switch and discharging the switch capacitance of the other the switch in the first switching leg via a resonant inductor connected between the junction points; and during the dead time when one of the switches in the second switching leg has been turned off, charging the switch capacitance of the turned-off switch and discharging the switch capacitance of the other the switch in the second switching leg via the resonant inductor connected between the junction points. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The foregoing aspects and the attendant advantages of the present invention will become more readily appreciated by reference to the following detailed description, when taken in conjunction with the accompanying drawings, wherein: 
       FIG. 1  shows two prior art circuits to illustrate the worst case scenario when a full bridge power supply is at no-load; 
       FIG. 1A  illustrate a waveform showing power loss versus load for a typical prior art ZVS full bridge converter; 
       FIG. 2  is a circuit diagram of a prior art full bridge power converter having a primary side resonant inductor for achieving soft, zero voltage switching; 
       FIG. 2A  is a set of voltage and current waveforms illustrating the operation of the power converter shown in  FIG. 2 ; 
       FIG. 3  is a prior art full bridge converter including two saturable inductors in series with secondary side rectifier diodes; 
       FIG. 4  shows a prior art full bridge converter including two external resonant inductors and two split capacitors for generating a split bulk voltage rail; 
       FIG. 4A  is a set of voltage and current waveforms illustrating the operation of the power converter in  FIG. 4 ; 
       FIG. 5  is a preferred embodiment of the circuit according to the present invention; 
       FIG. 5A  is an exemplary waveform of power loss versus load for the converter shown in  FIG. 5  as compared to the prior art converter shown in  FIG. 1 ; 
       FIG. 5B  is set of exemplary voltage and current waveforms for the circuit shown in  FIG. 5 ; 
       FIG. 6  is an alternative embodiment of the converter of the present invention that includes two resonant inductors; 
       FIG. 6A  shows a set of voltage and current waveforms and component values for the circuit shown in  FIG. 6 ; 
       FIG. 7  is an alternative embodiment of the circuit according to the present invention for use in a hard switching full bridge converter; 
       FIG. 7A  is a set of waveforms and component values for an exemplary converter in  FIG. 7 ; 
       FIG. 8  is an embodiment of the circuit according to the present invention for use in a hard switching half bridge converter, and 
       FIG. 9  is an alternative embodiment of the circuit in  FIG. 5  wherein the resonant inductor Lr is integrated in the power transformer as a magnetizing inductance by gapping the core. 
   

   Reference symbols or names are used in the Figures to indicate certain components, aspects or features shown therein, with reference symbols common to more than one Figure indicating like components, aspects or features shown therein. 
   DETAILED DESCRIPTION OF THE INVENTION 
   A preferred embodiment of a circuit according to the present invention is shown at  100  in  FIG. 5 . Converter  100  includes a resonant inductor Lr and a blocking inductor Lblock as seen in  FIG. 5 . Resonant inductor Lr is connected across the output terminals of a bridge  110 . Inductor Lr is parallel to the series combination of the transformer primary and a blocking inductor Lblock. A simplified representation of switches Qa, Qb, Qc, and Qd is shown in  FIGS. 5–8  such that the switch capacitances of the corresponding switches are not shown. The existence of the switch capacitances is well known in the art. For reference, the switch capacitances are as shown in bridge  20  for switches Qa, Qb, Qc, and Qd in  FIG. 1 . 
   In operation, when a bridge diagonal, e.g., Qa, Qd, activates, current builds up through the primary winding after overcoming the blocking period of Lblock. At the same time, current builds up in Lr in the same direction, while storing energy therein. At the end of the active period when the switch Qd turns off, the energy in inductor Lr is used to charge the switch capacitance of switch Qd, e.g., Cd (not shown) and discharge the switch capacitance of switch Qc, e.g., Cc (not shown) during the active to passive state transition. The bridge switches are preferably MOSFETs as shown in  FIG. 5 . Just before the next diagonal conduction, switch Qa turns off and the voltage at the top end of switch Qb starts to fall due to current flowing in inductor Lr. This current flow provides the charging of switch Qa and the discharging of switch Qb for the ZVS transition. For the converter in  FIG. 5 , if Lblock was missing, there is a potential for problems in the passive to active leg transition. As the upper end of switch Qb starts to fall, the current in inductor Lr may just circulate in the transformer primary, resulting in partial hard switching of this leg. Lblock provides a block to quickly prevent this reversal of current and the energy in inductor Lr is available for the ZVS transition. Similarly, in the next diagonal operation of switch Qc and switch Qb, current builds up in the primary and inductor Lr in the other direction and the same ZVS action described above occurs. 
   In  FIG. 5 , Lblock is chosen to act like a switch that would close after a short delay after reversal of voltage across it. Lblock could be a saturable choke or any kind of inductor which offers high impedance to a sudden reversal of current. It is not required nor expected that inductor Lblock stores energy. In an alternative embodiment, for particular applications where the transformer has a high enough leakage inductance, Lblock could be omitted. 
   An alternative embodiment wherein the resonant inductor Lr is integrated in the power transformer as a magnetizing inductance, Lmag, by gapping the core, is shown at  500  in  FIG. 9 . 
     FIG. 5B  is set of exemplary voltage and current waveforms for the circuit in  FIG. 5 . As seen in  FIG. 5B , unlike the “near square wave” current in inductor Lr in the waveform in  FIG. 2A  for the converter in  FIG. 2 , the current in inductor Lr for the converter in  FIG. 5  is triangular. Thus, for the converter in  FIG. 5 , the inductor Lr handles much lower current, resulting in lower copper and core losses as compared to Lr in  FIG. 2 . Inductor Lr in  FIG. 5  can therefore be a low cost gapped ferrite inductor, although other suitable energy storage materials can also be used. 
   Since the energy stored in inductor Lr is independent of load, ZVS action is achieved even at no-load condition. The converter in  FIG. 5  provides significant improvement in losses at light load of high frequency ZVS converters, as illustrated in  FIG. 5A .  FIG. 5A  is an exemplary waveform of power loss versus load for the converter shown in  FIG. 5 .  FIG. 5A  illustrates that, in contrast to the bridge devices of the conventional ZVS converter shown in  FIG. 1 , which has high losses at light loads, a substantial improvement at light load and no load is attained according to the present invention. As seen in  FIG. 5A , the losses at full load for the converter in  FIG. 5  rise marginally, but this is acceptable for most applications. These extra losses at full load for the embodiment shown in  FIG. 5  are due to higher currents seen when switching at full load. Alternatively, the circuit of  FIG. 5  can be optimized for a particular application by tuning inductor Lr to eliminate the extra loss at full load while allowing marginally higher losses at light load where such losses can be tolerated for the particular application. 
   As mentioned above, the prior art converter shown in  FIG. 4  may provide zero voltage switching down to very light loads for all four full bridge MOSFETS, Qa, Qb, Qc and Qd, if the power transformer is non-ideal, i.e., has high leakage inductance. As compared to the converter in  FIG. 4  with a non-ideal transformer, however, the present invention has the advantage of significantly reduced component cost and component count. If the power transformer has very low leakage inductance, then the converter in  FIG. 4  can achieve zero voltage switching of the two switches, which turn on after the transformer active period, i.e., the active to passive leg. The other leg, the passive to active leg, will not achieve ZVS action since the energy stored in the ZVS choke would find a discharge path through the power transformer to the secondary side load. In further contrast to the converter of  FIG. 4 , the series blocking inductance Lblock according to the present invention can provide blocking to achieve the ZVS action for all of the switches. 
     FIG. 6  is an alternative embodiment of the converter of the present invention that includes two resonant inductors. Power converter  200  includes an inductor Lr 1  connected in series with a diode D 3  between the bridge outputs, and an inductor Lr 2  connected in series with a diode D 4  between the bridge outputs. Each of the inductors Lr 1 , Lr 2  operates in a discontinuous mode and for only half the switching period.  FIG. 6A  shows a set of waveforms and component values for an exemplary circuit shown in  FIG. 6 . In operation, when the diagonal switches Qa–Qd in  FIG. 6  are on, current builds up in inductor Lr 2 , through diode D 4 , due to the voltage applied across inductor Lr 2 . As a result, energy is stored in the inductor Lr 2 . When switches Qa and Qd are turned off, this energy in inductor Lr 2  is used for the ZVS action of the other two switches, Qb and Qc. When the diagonal Qb–Qc is on, the same action occurs using inductor Lr 1  and diode D 3 . The alternative embodiment in  FIG. 6  provides a choice of splitting the heat dissipation of inductor Lr into two inductors in order to achieve the spreading of heat dissipation, a design choice which may be desirable in order to meet certain packaging conditions. 
   The circuit of the present invention may also be used in a soft switched half bridge converter. 
     FIG. 7  is an alternative embodiment of the circuit according to present invention for use in a hard switching full bridge converter. For a hard switched full bridge converter  300  in  FIG. 7 , instead of a phase shifted control circuit as used with the circuit in  FIG. 5 , Qa and Qd are controlled by the same drive signal, DrvA, from a conventional control circuit (not shown) so that these switches turn on and turn off at the same time. After these switches turn off, switches Qc and Qb are turned on at the same time by the same drive signal, DrvB, from a conventional control circuit (not shown). Both of these switch diagonals have the same on period and the pulse width is controlled to regulate the output voltage. Typically, the duty cycle for the hard switched full bridge converter is always less than 50%. In a particular application, if the duty cycle is very close to 50%, e.g., more than about 45%, the circuit of the present invention can be used in a hard switched converter, as seen in  FIG. 7 , for achieving ZVS.  FIG. 7A  is a set of waveforms and component values for an exemplary converter in  FIG. 7  operating at a 44% duty cycle. Thus, as is seen, the present invention can be used to overcome the basic drawback of hard switching and thereby achieve soft switching. 
     FIG. 8  is an embodiment of the circuit according to present invention for use in a hard switching half bridge converter. Thus, the present invention can be used to enable a low cost, simple hard switching full bridge or half bridge converter to achieve zero voltage switching, if the operating duty cycle is close to 50%. 
   As described above, the present invention achieves zero voltage switching of full bridge or half bridge devices even at very light loads, solves the light load power loss issue in soft switching full bridge and half bridge converters operating at high operating frequencies, and reduces cost by enabling use of lower cost components. The present invention can be used in soft switched full bridge as well as half bridge converters. The present invention can also be used to get zero voltage transition switching in hard switched bridge topologies where the operating duty cycle is very large with very short dead time. 
   Having disclosed exemplary embodiments, modifications and variations may be made to the disclosed embodiments while remaining within the scope of the invention as described by the following claims.