Abstract:
A feedback loop is used to optimize a zero current threshold for a switching regulator. After the low side power switch of the switching regulator turns off, the switching node state is monitored to adjust the zero current threshold in a real time and thus the low-side power switch is prevented from turning off too early or too late. Thereby the efficiency in green mode is optimized.

Description:
FIELD OF THE INVENTION 
     The present invention is related generally to switching regulators and, more particularly, to zero current detection for switching regulators. 
     BACKGROUND OF THE INVENTION 
     In green mode, for example, pulse skipping mode, diode emulated mode, etc, a switching regulator will detect zero current for disabling the low-side power switch to prevent from reverse inductor current that may cause extra efficiency loss. Traditionally, the zero current detection is implemented by comparing the voltage of the switching node with some threshold such as ground potential, for example, disclosed by U.S. Pat. No. 7,327,127. However, the threshold for identifying a zero current is fixed in the circuit design and thus, when some component has different characteristics, for example, the comparator offset and delay time, the resistance of the high-side or low-side power switch, and the noise effect at the switching node, the low-side power switch may be turned off too early or too late, and consequently it can&#39;t precise to optimize the efficiency in green mode. 
     SUMMARY OF THE INVENTION 
     An objective of the present invention is to provide a real time adjustable zero current detector and detection method for a switching regulator. 
     According to the present invention, a real time adjustable zero current detector for a switching regulator uses a feedback loop to monitor the switching node state after a low-side power switch turns off to optimize a zero current threshold, and a comparator to compare the voltage of the switching node with the zero current threshold to trigger a zero current signal. 
     According to the present invention, a real time adjustable zero current detection method for a switching regulator monitors the switching node state after a low-side power switch turns off to optimize a zero current threshold, and compares the voltage of the switching node with the zero current threshold to trigger a zero current signal. 
     By monitoring the switching node state to adjust the zero current threshold in a real time, the low-side power switch is prevented from turning off too early or too late, thereby optimizing the efficiency in green mode. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       These and other objectives, features and advantages of the present invention will become apparent to those skilled in the art upon consideration of the following description of the preferred embodiments of the present invention taken in conjunction with the accompanying drawings, in which: 
         FIG. 1  is a circuit diagram of a first embodiment according to the present invention; 
         FIG. 2  is a waveform diagram of the switching regulator shown in  FIG. 1 ; 
         FIGS. 3A-3C  illustrates an embodiment for identifying an up signal and a down signal with two sample values; 
         FIGS. 4A-4C  illustrates an embodiment for identifying an up signal and a down signal with three sample values; 
         FIG. 5  is a circuit diagram of a second embodiment according to the present invention; 
         FIG. 6  is a waveform diagram of the switching regulator shown in  FIG. 5 ; 
         FIG. 7  is a circuit diagram of a third embodiment according to the present invention; 
         FIG. 8  is a waveform diagram of the switching regulator shown in  FIG. 7 ; 
         FIG. 9  is a circuit diagram of a fourth embodiment according to the present invention; and 
         FIG. 10  is a waveform diagram of the switching regulator shown in  FIG. 9 . 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       FIG. 1  is a circuit diagram of a first embodiment according to the present invention, in which a switching regulator includes a high-side power switch M 1  connected to a low-side power switch M 2  through a switching node  10 , a controller chip  12  to provide control signals Vug and Vlg to switch the high-side power switch M 1  and the low-side power switch M 2  to generate an inductor current IL to charge a capacitor CL to thereby generate an output voltage Vo, divider resistors R 1  and R 2  to divide the output voltage Vo to generate a feedback signal VFB for the controller chip  12 , and a compensation network  14  to compensate the feedback signal VFB. In the controller chip  12 , a pulse width modulation (PWM) control logic  16  provides a pulse width modulation signal PWM for drivers  18  and  20  to generate the control signals Vug and Vlg, and a zero current detector  22  includes a feedback loop  24  and a comparator  26  for identifying a zero current. The feedback loop  24  monitors the switching node state after the low-side power switch M 2  turns off to optimize a zero current threshold Vzc, and the comparator  26  compares the voltage Vx of the switching node  10  with the zero current threshold Vzc to trigger a zero current signal ZC for the driver  20  to disable the low-side power switch M 2 . In the feedback loop  24 , a comparator  28  compare the voltage Vx with a reference voltage Vr to generate a comparison signal Sc, a timing generation circuit  30  detects the control signal Vlg to trigger a timing signal Sp when the low-side power switch M 2  turns off, for a control logic  32  to sample the comparison signal Sc to determine an up signal U and a down signal D, an N-bit up/down counter  34  increases or decreases a count value CNT depending on the up signal U and the down signal D, a digital-to-analog converter (DAC)  36  converts the count value CNT into an analog voltage Vzc′, and a buffer  38  generates the zero current threshold Vzc from the analog voltage Vzc′. 
       FIG. 2  is a waveform diagram of the switching regulator shown in  FIG. 1 . Referring to  FIG. 1  and  FIG. 2 , as shown by waveform  40  at time t 1 , when the voltage Vx increases to the zero current threshold Vzc, the control signal Vlg turns to low, as shown by waveform  42 , which will turn off the low-side power switch M 2  and make the timing generation circuit  30  to trigger the timing signal Sp, as shown by waveform  44 , to signal the control logic  32  to sample the comparison signal Sc at time t 2 . At this time t 2 , since the voltage Vx is higher than the reference voltage Vr, as shown by waveform  40 , the sample value will be “1”, leading the up signal U to high and the down signal D to low. As a result, the count value CNT increase to “01110” from “01101”, as shown by a bar  46 , and thereby decreases the zero current threshold Vzc. On the contrary, when the sample value is “0”, as shown by time t 3 , the up signal U is low and the down signal D is high, resulting in the count value CNT decreasing to “01101” from “01110”, thereby increasing the zero current threshold Vzc. 
     In the embodiment shown in  FIG. 2 , the control logic  32  determines the up signal U and the down signal D with only a sample. However, in other embodiments, more samples may be taken into consideration to determine the up signal U and the down signal D. For example,  FIG. 3  shows the control logic  32  determines the up signal U and the down signal D with two sample values. In  FIG. 3A , the values of the voltage Vx at two sampling time points SH 1  and SH 2  are both lower than the reference voltage Vr, so the control logic  32  obtains the sampled result of “00”, which makes the up signal U and the down signal D both at high, thereby increasing the zero current threshold Vzc. In  FIG. 3B , the value of the voltage Vx at the first sampling time SH 1  is lower than the reference voltage Vr, while the value of the voltage Vx at the second sampling time SH 2  is higher than the reference voltage Vr, so the control logic  32  obtains the sampled result of “01”, which makes both the up signal U and the down signal D at low, thereby keeping the zero current threshold Vzc unchanged. In  FIG. 3C , the values of the voltage Vx at two sampling time points SH 1  and SH 2  are both higher than the reference voltage Vr, so the control logic  32  obtains the sampled result of “11”, which makes the up signal U at high and the down signal D at low, thereby decreasing the zero current threshold Vzc. 
     Alternatively,  FIG. 4  illustrates another embodiment that the control logic  32  determines the up signal U and the down signal D with three sample values. In  FIG. 4A , the values of the voltage Vx at three sampling time points SH 1 , SH 2  and SH 3  are all lower than the reference voltage Vr, so the control logic  32  obtains the sampled result of “000”, thereby making the up signal U at low and the down signal D at high, which will increase the zero current threshold Vzc. In  FIG. 4B , the values of the voltage Vx at the first and second sampling time points SH 1  and SH 2  are lower than the reference voltage Vr, and the voltage Vx at the third sampling time point SH 3  is higher than the reference voltage Vr, so the control logic  32  obtains the sampled result of “001”, thereby making the up signal U and down signal D both at low, which will remain the zero current threshold Vzc. In  FIG. 4C , the values of the voltage Vx at three sampling time points SH 1 , SH 2  and SH 3  are all higher than the reference voltage Vr, so the control logic  32  obtains the sampled result of “111”, thereby making the up signal U at high and the down signal D at low, which will decrease the zero current threshold Vzc. 
       FIG. 5  is a circuit diagram of a second embodiment according to the present invention, which is similar to that shown in  FIG. 1  except the circuit in the feedback loop  24 . In  FIG. 5 , the timing generation circuit  30  detects the control signal Vlg to trigger the timing signal Sp when the low-side power switch M 2  turns off, for a sample-and-hold circuit  48  to sample the voltage Vx to generate a sample signal LXF, a comparator  28  compares the sample signal LXF with the reference voltage Vr to generate the comparison signal Sc, and the N-bit up/down counter  34  increases or decreases the count value CNT depending on the comparison signal Sc. The rest part of this embodiment is the same as that of  FIG. 1 , in terms of both configuration and operation. 
       FIG. 6  is a waveform diagram of the switching regulator shown in  FIG. 5 . Referring to  FIG. 5  and  FIG. 6 , as shown by waveform  40  at time t 4 , when the voltage Vx increases to the zero current threshold Vzc, the control signal Vlg turns to low, as shown by waveform  42 , and thus turns off the low-side power switch M 2  and signal the timing generation circuit  30  to trigger the timing signal Sp, as shown by waveform  44 . Consequently, the sample-and-hold circuit  48  samples the voltage Vx at time t 5 . At this time t 5 , since the voltage Vx is higher than the reference voltage Vr, as shown by waveform  40 , the resultant sample signal LXF will be also higher than the reference voltage Vr, as shown by waveform  50 , resulting in the output Sc of the comparator  28  at high. Therefore, the N-bit up/down counter  34  increases the count value CNT from “01101” to “01110”, as shown by a bar  46 , which will decrease the zero current threshold Vzc. On the contrary, when the sample signal LXF is lower than the reference voltage Vr, as shown at time t 6 , the output Sc of the comparator  28  will be low, and thus the N-bit up/down counter  34  will decrease the count value CNT from “01110” to “01101”, thereby increasing the zero current threshold Vzc. 
       FIG. 7  is a circuit diagram of a third embodiment according to the present invention, which is modified by replacing the digital circuit that generates the analog voltage Vzc′ of  FIG. 1  by an analog circuit. In this embodiment, the up signal U and the down signal L provided by the control logic  32  control a charge/discharge circuit  52  to charge or discharge a capacitor Czc to generate the analog voltage Vzc′. In the charge/discharge circuit  52 , a first current source  54  determines a charge current Iu, a charge switch SW 1  is connected between the first current source  54  and the capacitor Czc, a second current source  56  determines a discharge current Id, and a discharge switch SW 2  is connected between the capacitor Czc and the second current source  56 . When the up signal U is high and the down signal L is low, the charge switch SW 1  is on and the discharge switch SW 2  is off, so that the charge current Iu charges the capacitor Czc, thereby increasing the analog voltage Vzc′. When the down signal L is high and the up signal U is low, the charge switch SW 1  is off and the discharge switch SW 2  is on, so that the discharge current Id discharges the capacitor Czc, thereby decreasing the analog voltage Vzc′. 
       FIG. 8  is a waveform diagram of the switching regulator shown in  FIG. 7 . Referring to  FIG. 7  and  FIG. 8 , as shown by waveform  40  at time t 7 , when the voltage Vx increases to the zero current threshold Vzc, the control signal Vlg turns to low, as shown by waveform  42 , so the low-side power switch M 2  is turned off and the timing generation circuit  30  triggers the timing signal Sp, as shown by waveform  44 . Consequently, the control logic  32  samples the comparison signal Sc at time t 8 . At the time of sampling, if the voltage Vx is higher than the reference voltage Vr, as shown by waveform  40 , the sampled result will be “1”, and the control logic  32  triggers the up signal U that has a constant pulse width, thereby turning on the charge switch SW 1  to charge the capacitor Czc to increase the analog voltage Vzc′, as shown by waveform  58 , which will decrease the zero current threshold Vzc. On the contrary, if the voltage Vx is lower than the reference voltage Vr when the control logic  32  performs sampling, as shown by waveform  40  at time t 9 , the sampled result will be “0”, and the control logic  32  triggers the down signal L that has a constant pulse width, thereby turning on the discharge switch SW 2  to discharge the capacitor Czc to decrease the analog voltage Vzc′, as shown by waveform  58 , which will increase the zero current threshold Vzc. In this embodiment, the control logic  32  determines the up signal U and the down signal L with only a sample. However, in other embodiments, more samples may be taken into consideration to determine the up signal U and the down signal L, as illustrated in  FIG. 3  and  FIG. 4 . 
       FIG. 9  is a circuit diagram of a fourth embodiment according to the present invention, which is modified by replacing the digital circuit that generates the analog voltage Vzc′ of  FIG. 5  by an analog circuit. In this embodiment, the comparator  28  compares the sample signal LXF with the reference voltage Vr to generate a pair of complementary first and second comparison signals Scu and Scd to control the charge/discharge circuit  52  to charge or discharge the capacitor Czc to generate the analog voltage Vzc′. When the first comparison signal Scu is high, the comparison signal Scd is low, so the charge switch SW 1  is on and the discharge switch SW 2  is off, resulting in the charge current Iu to charge the capacitor Czc to increase the analog voltage Vzc′. When the second comparison signal Scd is high, the first comparison signal Scu is low, so the charge switch SW 1  is off and the discharge switch SW 2  is on, resulting in the discharge current Id to discharge the capacitor Czc to decrease the analog voltage Vzc′. 
       FIG. 10  is a waveform diagram of the switching regulator shown in  FIG. 9 . Referring to  FIG. 9  and  FIG. 10 , as shown by waveform  40  at time t 10 , when the voltage Vx increase to the zero current threshold Vzc, the control signal Vlg turns to low, as shown by waveform  42 . Thus, the low-side power switch M 2  is turned off and the timing generation circuit  30  triggers the timing signal Sp, as shown by waveform  44 , so that the sample-and-hold circuit  48  samples the voltage Vx at time t 11  and generates the sample signal LXF. If the sample signal LXF is higher than the reference voltage Vr, as shown by waveform  50  at time t 11 , the comparator  28  triggers the comparison signal Scu with a constant pulse width to turn on the charge switch SW 1  to charge the capacitor Czc, causing the analog voltage Vzc′ to increase, as shown by waveform  58 , and in turn decreasing the zero current threshold Vzc. On the contrary, if the sample signal LXF is lower than the reference voltage Vr, as shown by waveform  50  at time t 12 , the comparator  28  triggers the comparison signal Scd with a constant pulse width to turn on the discharge switch SW 2  to discharge the capacitor Czc, causing the analog voltage Vzc′ to decrease, as shown by waveform  58 , and in turn increasing the zero current threshold Vzc. 
     While the present invention has been described in conjunction with preferred embodiments thereof, it is evident that many alternatives, modifications and variations will be apparent to those skilled in the art. Accordingly, it is intended to embrace all such alternatives, modifications and variations that fall within the spirit and scope thereof as set forth in the appended claims.