Abstract:
A network for generating a set of intermediate voltages comprising two input ports for feeding two reference voltages. The intermediate voltages are generated by a number of self calibration units that correspond to the number of intermediate voltages to be generated. Each self calibration unit receives the voltages of the neighboring calibration units or the voltage of one neighboring calibration unit and one of the reference voltages.

Description:
BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The invention relates to a circuit and a method for generating a set of, for example, linearly or logarithmically spaced intermediate voltages as needed for analog-to-digital conversion, particularly with flash ADC converters. 
   2. Related Technology 
   Several types of analog-to-digital converters (ADC), notably flash and folding converters, operate by comparing an input voltage with a set of reference voltages uniformly distributed over the input signal full scale. The accuracy of these reference voltages is a key factor determining the linearity of the ADC. The present disclosure describes how a set of linearly spaced voltages can be produced as the collective result of an array of interacting self-calibration units. The invention lends itself well to a high-accuracy implementation ensuring linear spacing without resorting to calibration on an external reference. With a minor change in the design of the self-calibration unit, the array can also produce logarithmically spaced reference voltages. 
   By far the most common way to generate a set of linearly spaced voltages consists of using a chain of identical resistors as shown, for example, in U.S. Pat. No. 6,437,724 B1. When the smallest desired voltage is applied to one end of the chain and the largest desired voltage is applied to the other end, intermediate taps in the chain settle to intermediate voltages with uniform spacing. This is illustrated in  FIG. 1 . 
   This approach, while sufficient for many applications, suffers from some drawbacks: 
   Accuracy is limited by resistor matching. Up to a point, matching can be improved by increasing the geometrical size of each resistor, but for highest accuracy, calibration becomes necessary. 
   When a current is drawn from the taps of the resistor chain—such as input bias currents of comparators for instance—the tap voltages are no longer uniformly spaced. In order to reduce the impact of such parasitic currents, it is generally necessary to choose very small resistor values, which results in a large power dissipation in the resistor chain. 
   The output impedance is not the same for all taps of the resistor chain. This drawback is significant in the case of fully differential ADC architectures, where the voltage boundaries are not constant but consist of the input signal of the ADC. At high frequencies, different taps will have different bandwidths and group delays, which creates distortion. 
   SUMMARY OF THE INVENTION 
   The invention provides a circuit and a method for generating a set of spaced voltages wherein the voltages are not dependent on properties of electrical devices, power dissipation is minimized, and well defined and constant output characteristics are guaranteed. 
   The invention provides a circuit and a method for generating a set of intermediate voltages. 
   According to the invention the circuit for generating a set of intermediate voltages includes two input ports for feeding two reference voltages, wherein the intermediate voltages are generated by a number of self calibration units that correspond to the number of intermediate voltages to be generated, wherein each self calibration unit receives the voltages of the neighboring calibration units or the voltage of one neighboring calibration unit and one of the reference voltages. 
   In a preferred embodiment, each self calibration unit includes an error amplifier network or circuit for providing an error voltage, a comparator for providing an up/down signal when the error voltage exceeds a positive or a negative voltage level, an up/down counter serving as an integrator and counting an internal count one up or one down depending on the up/down signal received from the comparator, an digital-analog-converter for converting the digital signal received from the up/down counter into an analog signal, and a clock signal generator for providing clock signals for the switches and the up/down counter. 
   In particular, the error amplifier network or circuit preferably includes a first capacitor which receives an output voltage V k  of the self-calibration unit via a first switch and receives the output voltage V k−1  of a first neighboring self-calibration unit via a second switch, a second capacitor which receives an output voltage V k  of the self-calibration unit via a third switch and the output voltage V k+1  of a second neighboring self-calibration unit via a fourth switch, an operational amplifier wherein the first and second capacitor are connected to its negative input port and its positive input port is connected to earth potential, moreover, a feedback capacitor which is connected with one terminal to the negative input port of the operational amplifier and with the other terminal to the output port of the operational amplifier. 
   In a preferred embodiment, each capacitor can be switched alternatively towards the V k+1  input and the V k−1  input instead of always going to the same input. Thus, on the average, the amplifier network or circuit will have the same gain for both inputs even if the two capacitors do not have exactly the same value. 
   In another preferred embodiment, each self-calibration unit tends to make V k+1 −V k  twice as large as V k −V k−1 . Then, the result of an array of such units is a set of voltage differences in geometric progression with a radix of two. 
   In the general case, the self-calibration units are preferably designed in such a way so that for the voltage differences the following equation is satisfied:
 
α·( V   k+1   −V   k )=(1−α)·( V   k   −V   k−1 ), 0&lt;α&lt;1.
 
   When α=½, such a self-calibration will produce linearly spaced voltages. When α=⅓, it produces voltages in a geometric progression with a radix of 2. When α=1/(1+N), the radix of the geometric progression is N. 
   In one alternative embodiment, the self-calibration units are chained to a binary tree, wherein one unit takes two boundary voltages as inputs and produces a first middle voltage and two other units determine the middle point between one boundary voltage and the first middle voltage and the middle point between the first middle voltage and the other boundary voltage. Thus, adding layer after layer of additional units can be added to find the middle points within the set of voltages produced by previous layers generating a binary tree of voltages. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     Embodiments of the invention will now be described in more detail with reference to the drawings. In the drawings, 
       FIG. 1  shows a prior art circuit using a resistor chain to produce four linearly spaced reference voltages between two boundary voltages; 
       FIG. 2  shows an embodiment of the invention producing four evenly spaced reference voltages between two boundary voltages; 
       FIG. 3  shows simulated voltages versus iteration time starting from a random initial pattern; 
       FIG. 4  shows an embodiment of a self-calibration unit based on a switched-capacitor error amplifier and a digital up/down counter serving as an integrator; 
       FIG. 5  shows another embodiment of the invention producing four evenly spaced reference voltages between two boundary voltages; and 
       FIG. 6  shows another embodiment of a self-calibration unit. 
   

   DETAILED DESCRIPTION 
   A block diagram of an embodiment of the invention producing a total of six linearly spaced voltages, including boundary voltages, is shown in  FIG. 2 . The boundary voltages V 0  and V 5  are produced by constant sources  2 , 3 . An array of four self-calibration units  100  produces the inner four voltages V 1  through V 4 . Each self-calibration unit  100  has two inputs connected to the outputs of its two nearest neighbours in the array. 
   At all times, each self-calibration unit  100  regulates its own output voltage toward the middle point between its two input voltages (V 0 , V 2 ), (V 1 , V 3 ), (V 2 , V 4 ), (V 3 , V 5 ). In response to a voltage change in any unit  100 , the neighbours will respond by correcting their own output voltages in an attempt to re-centre it. Starting from any initial voltage pattern, the array as a whole will converge toward a state where all voltages are uniformly spaced between V 0  and V 5 , in the sequence defined by the topology of the array. In mathematical terms, the self-calibration unit controlling voltage V k  computes an error voltage
 
 E   k ( t )=( V   k ( t )− V   k−1 ( t ))−( V   k+1 ( t )− V   k ( t ))=2 V   k ( t )− V   k−1 ( t )− V   k+1 ( t )  (1)
 
   and permanently updates V k  in such a way to reduce the error E k . The simplest embodiment of a calibration unit  100  is a first-order integrator. In continuous time, the operation of a self-calibration unit  100  can thus be described by the following differential equation: 
   
     
       
         
           
             
               
                 
                   
                     ⅆ 
                     
                       V 
                       k 
                     
                   
                   
                     ⅆ 
                     t 
                   
                 
                 = 
                 
                   
                     - 
                     τ 
                   
                   · 
                   
                     
                       E 
                       k 
                     
                     ⁡ 
                     
                       ( 
                       t 
                       ) 
                     
                   
                 
               
             
             
               
                 ( 
                 2 
                 ) 
               
             
           
         
       
     
   
   where τ is the time constant of the integrator. In some embodiments, the self-calibration units  100  may operate in discrete time, in which case they update their output voltage at specific time points determined by a clock signal. 
   The invention readily generalizes to arrays of any size. A discrete-time simulation in the case of an array of nine units is shown in  FIG. 3 . 
     FIG. 4  shows an embodiment of the self-calibration unit  100 . This circuit includes an error amplifier, comprising elements  101 - 109 , determining the direction in which the next correction must be made, and a digital up/down counter  113  serving as an integrator. The error amplifier applies switched-capacitor techniques in order to achieve high accuracy despite the amplifier offset. 
   In  FIG. 4 , circles containing a number represent switches  101 - 105  controlled by the clock generator unit  112 . The number indicates in which clock phase the switch is closed. In phase  1 , both input capacitors  106 ,  107  are connected to the output node to receive the output voltage V k  of this actual self-calibration unit, whereas the amplifier  109  is configured as a voltage follower. During this phase, both input capacitors  106 ,  107  of capacitance C are pre-charged to voltage V k , minus the amplifier offset. In phase  2 , the input capacitors  106 ,  107  are switched over to inputs V k−1  and V k+1 . For one of the capacitors  106 , this change produces a positive voltage step whereas for the other one  107 , it produces a negative voltage step. 
   The amplifier offset remains present at its inverting input. If V k  is exactly in the middle between V k−1  and V k+1 , the net effect of both steps is exactly zero, therefore the output of the amplifier will not change. Any deviation of V k  from the middle point will cause a net amount of charge to flow onto feedback capacitor C fb . The voltage error appears at the amplifier output, magnified by the capacitance ratio C/C fb . 
   A comparator  110  determines whether the error is positive or negative. The comparator output determines whether the counter  113  will increment or decrement its state at the next clock cycle. Hence, the counter output is the integral over time of the sign of the voltage error. A digital-to-analog converter (DAC)  114  turns the counter output into output voltage V k . The output voltage will converge toward a limit cycle around the middle point between V k+1  and V k−1 . The residual error can be made as small as necessary by increasing the resolution of the DAC  114 . The limit cycle could be avoided by using two comparators with slightly different thresholds so that the counter neither increments, nor decrements when the residual error is very small. 
   The advantage of a digital integrator is that its state will remain preserved for as long as the circuit is powered up, even if the self-calibration process is stopped. By contrast, analog integrators are typically affected by leakage currents causing the output voltage to drift, unless the self-calibration process is running permanently. 
   As described above, the self-calibration circuit  100  relies on good matching between the two input capacitors  106 ,  107  of nominal value C. The circuit of  FIG. 4  can be further refined in order to provide dynamic matching between these two capacitors. At the cost of a few more switches, each capacitor can be switched alternatively toward the V k+1  input and the V k−1  input instead of always going to the same input. On average, the circuit  100  will have the same gain for both inputs even if the two capacitors  106 ,  107  do not have exactly the same value. 
   Instead of linearly spaced voltages, it is possible to obtain logarithmically spaced voltages by a minor change in the design of the self-calibration unit  100 . Instead of aiming for the middle point between V k−1  and V k+1 , the unit can tune V k  to split the interval from V k−1  to V k+1  in another ratio than unity. For instance, if each unit tends to make V k+1 −V k  twice as large as V k −V k−1 , then the collective result of an array of such units is a set of voltage differences in geometric progression with a radix of two. Unequal splitting ratios can be produced by assigning different values to the two input capacitors. 
   A progression with a radix of two is only one particular case. The general case includes building a self-calibration unit in such a way that they tend to produce the following relationship between node voltages:
 
α·( V   k+1   −V   k )=(1−α)·( V   k   −V   k−1 ), 0&lt;α&lt;1
 
   When α=½, such a self-calibration will produce linearly spaced voltages. When α=⅓, it produces voltages in a geometric progression with a radix of 2. When α=1/(1+N), the radix of the geometric progression is N. 
   The only network topology described above includes chaining self-calibration units  100  in a one-dimensional string. Other network topologies are possible, which may have somewhat different properties in terms of robustness to imperfections in the self-calibration units  100 . One alternative example consists of a binary tree. One unit may take the two boundary voltages as inputs and adjust itself to produce the middle voltage of the range. Two other units may determine the middle point between one boundary voltage and the middle voltage determined by the first unit. Layer after layer of additional units could be added to find the middle points within the set of voltages produced by the previous layer. This topology may offer better integral non-linearity in the presence of imbalance within the self-calibration units, possibly at the cost of some degradation in differential non-linearity. 
   Subsequently, a more detailed description of the self-calibration unit in  FIG. 4  is provided. A clock generator  112  provides at its output a clock signal which is provided to switches  101 ,  102 ,  103 ,  104 , and  105  as well as to the input of an up/down counter  113 . In a first phase of the clock signal the switches  102 ,  103  and  105  are closed. In a second phase of the clock signal the switches  101  and  104  are closed. 
   In a first phase, the voltage V k  which is provided from the output of a digital-analog-converter  114  is fed via switches  102  and  103  to the first input terminals of capacitors  106  and  107  which have a first and a second capacitance C whereby switches  102  and  103  are closed. Second terminals of capacitors  106  and  107  are connected to a negative input port of feedback voltage supplying amplifier  109 . The positive input port of the feedback voltage supplying amplifier  109  is connected to earth potential. 
   The voltage V k , which is provided from the output of the digital-analog-converter  114 , charges the first and the second capacitor  106  and  107 . In the first phase the switch  105  is also closed and short-circuits a feedback capacitor  108  which has a capacitance of C fb . 
   In a second phase the switches  102  and  103  are open and the switches  101  and  104  are closed. In this phase the voltage V k−1  is connected via a switch  104  to the first input port of the second capacitor  107  and the voltage V k+1  is connected via a switch  101  to the first input port of the first capacitor  106 . The second terminals of capacitors  106  and  107  are connected to the first input port of the feedback capacitor  108  and the negative input port (−) of the feedback voltage supplying amplifier  109 . The second terminal of the feedback capacitor  108  is connected to the negative input port (−) of the comparator  110 . The positive input port (+) of the comparator  110  is connected to earth potential. In the second phase the switch  105  is open and, hence, feedback capacitor  108  is charged. 
   The voltage step from V k  to V k+1  at the first input port of capacitor  106  that happens when the capacitor charged in a first phase to voltage V k  is then switched in a second phase to V k+1  and introduces a first current in a first direction via first capacitor  106  and feedback capacitor  108 . Conversely, the voltage step from V k  to V k−1  at the first input port of capacitor  107  that happens when the capacitor charged in a first phase to voltage V k  is then switched in a second phase to V k−1  introduces a second current in a second direction via second capacitor  107  and feedback capacitor  108 . 
   If the voltage V k  in the first phase is exactly the arithmetic mean of the voltages V k+1  and V k−1  that are connected to the first input terminal of the first capacitor  106  and the first terminal of the second capacitor  107 , respectively, the net current over the feedback capacitor  108  is zero. If the voltage V k  in the first phase was not exactly the arithmetic mean of the voltages V k+1  and V k−1  that are connected to the first input terminal of the first capacitor  106  and the first terminal of the second capacitor  107 , respectively, the net current over the feedback capacitor  108  is not zero and the feedback capacitor  108  is charged. The charge of the feedback capacitor  108  is calculated as a difference voltage from the voltages V k+1 , V k−1 , and V k  and is a weighted average of the ratio of the capacitance of the first capacitor  106  and the feedback capacitor  108  and the capacitance of the second capacitor  107  and the feedback capacitor  108 . 
   The charge of the feedback capacitor  108  provides an error voltage to the negative input port of the comparator  110 . If the error voltage exceeds a preset positive voltage the comparator  110  will provide an up signal. If the error voltage exceeds a preset negative voltage the comparator  110  will provide a down signal. 
   The up or down signal of the comparator  110  is provided to the input port of the up/down counter  113 . The up/down counter  113  increments or decrements an internal count depending if an up signal or a down signal is provided at its input. The count of the up/down counter  113  is provided at its output to the input of the digital-analog-converter  114 , which converts the count of the up/down counter  113  to a voltage signal at its output port. The output of the digital-analog-converter  114  will provide the voltage V k  via the switches  102  and  103  to the first input terminals of the capacitors  106  and  107  and will charge the capacitors  106  and  107  in the first phase of the clock signal provided by the clock generator  112 . 
   As shown in  FIG. 5 , the invention is of particular interest in the case where each self-calibration unit  100  produces a voltage across an output resistor  1001 - 1004  connected to a common node. In this case, an input signal can be applied to the common node, whereby each output of the array tracks the input signal V in  with a constant offset, the offsets being linearly distributed over some range. Given that all outputs of the array track the same input signal simultaneously, voltage differences between array outputs are independent of the input signal. The self-calibration algorithm as described above will still produce the same result as if the common node was held at a constant voltage. 
   This variation is of particular interest because this structure is the basis of a particularly advantageous fully differential high-speed ADC front-end architecture as described in European Patent Application EP 05 019 801.9, which is incorporated into this application by reference. In this circuit, the signal paths from the common input to all output nodes V 0 -V 5  of the array have the same series impedance and loading conditions, which ensures that the bandwidth and propagation delays match between them. Resistor chains from the prior art can also be made to track an input signal, but in this case, the impedances vary widely between output taps V 0 -V 5 . 
   Subsequently, a second embodiment for producing a set of linearly spaced voltages is described. In this embodiment several resistors  1001 - 1004  are connected with their first port to a common node to which a voltage V in  can be connected. Current sources  1005 - 1008  are connected with their first port to the second port of resistors  1001 - 1004  and with their second port to earth potential. Current sources  1005 - 1008  are current sources with a third port for controlling their output current. 
   Also in this embodiment, the self-calibration units  100  receive input voltages at three ports. At a second port the self-calibration units  100  receive the voltage potentials V 1 -V 4  that can be tapped at the second ports of resistors  1001 - 1004 . At a first port the self-calibration units  100  receive the voltages V 1 -V 3  or a reference voltage V 0  in case of the left-most unit in  FIG. 5 . At a third port the self-calibration units  100  receive the voltages V 2 -V 4  or a reference voltage V 5  in case of the right-most unit in  FIG. 5 . 
     FIG. 6  shows another embodiment of the self-calibration unit  2000 . This embodiment comprises two summations  2001  and  2002 , two multiplier  2003  and  2004 , another summation  2005  and an integrator  2006 . An output voltage V k  is summed with input voltages V k+1  and V k−1  so that the difference voltages V k+1 −V k  and V k −V k−1  are obtained. These differences are scaled with factors α and 1−α so that the scaled differences α·(V k+1 −V k ) and (1−α)·(V k −V k−1 ) are obtained. These scaled differences are again summed and the error E k (t)=(1−α)·(V k −V k−1 )−α·(V k+1 −V k ) is obtained which is integrated to the output voltage V k . 
   The advantages of the previously described invention may be summarized as follows. 
   The accuracy of the set of voltages generated according to the invention can be higher than the accuracy achievable with resistor chains, because the self-calibration units  100  can be designed to rely on dynamic comparisons rather than static element matching. 
   Input bias currents possibly drawn by comparators loading the outputs do not distort the distribution of reference voltages produced by the invention, in contrast to resistor chains. 
   All voltage outputs are driven with the same impedance, which ensures identical bandwidths and propagation delays in variations where an input signal is tracked. 
   The invention is suitable as a self-calibration scheme for an ADC front-end circuit disclosed in EP 05 019 801.9. 
   The invention is not restricted to the above embodiments and can be used with different implementations. The current sources and the input stages of the differential amplifiers can be implemented in bipolar technology but also in CMOS technology. The current sources can be configured as current mirror circuits. All features described in this description and shown in the accompanying drawings can be combined.