Abstract:
A spectrum analyzer with a compensation circuitry for prevention of measurement accuracy deterioration due to local oscillators phase noise.

Description:
FIELD OF THE INVENTION 
   The present invention relates to spectrum analyzers that analyze the frequency spectrum of an incoming signal and display the spectrum on a frequency domain display. In particular, this invention relates to means for preventing measurement accuracy deterioration in spectrum analyzers that use local oscillators with substantial level of phase noise. 
   BACKGROUND OF THE INVENTION 
   A spectrum analyzer is a device that measures the power density of an input signal and displays that power density in a form convenient to the user. A typical block diagram of a prior art spectrum analyzer is shown in  FIG. 1 . The spectrum analyzer of  FIG. 1  includes a frequency converter  100 , a low pass filter (LPF)  101 , an analog-to-digital converter (ADC)  102 , a processor  103 , a display  104  and a control unit  105 . The frequency spectrum of an applied input signal is measured in a step-by-step process. The control unit  105  controls the frequency converter  100  and, in particular, specifies at each step, a frequency band Fst . . . Fst+ΔF of the input signal spectrum that is to be currently analyzed (Fst is the start frequency of the band to be analyzed, and ΔF is the spacing between adjacent start frequencies). The frequency converter  100  transfers the band Fst . . . Fst+ΔF to the band  0  . . . ΔF. An anti-aliasing low pass filter (LPF)  101  suppresses all components with the frequencies higher than Fs/2 (where Fs is the sampling rate). The analog to digital converter (ADC)  102  transforms an incoming continuous signal into a sequence of digital samples with the sampling rate Fs. The processor  103  carries out a Fast Fourier Transform of the signal that comes from the next frequency band at each next step of the spectrum measurement. Then, processor  103  concatenates the resulting partial spectrum pieces into an aggregate spectrum of the input signal and transfers the resulting spectrum to the display  104 , interacting all the time with the control unit  105 . 
   One of the essential conditions that should be met to achieve a high measurement accuracy in a spectrum analyzer, is a requirement for the frequency converter  100  not to create spurious responses, which may substantially distort the final picture. To attain such a purpose, a conventional frequency converter usually contains several conversion stages with an appropriate selection of intermediate frequencies and frequencies of local oscillators. As an example, a prior art spectrum analyzer with a three-stage frequency converter is shown in  FIG. 2 . 
   In the spectrum analyzer of  FIG. 2 , the first stage of the frequency converter  100  is formed by a first mixer  200 , a first band pass filter (BPF)  201  and a first local oscillator (LO)  211 . The second stage of frequency converter  100  is formed by a second mixer  202 , a second band pass filter  203  and a second local oscillator  212 . The third stage of frequency converter  100  is formed by a third mixer  205  and a third local oscillator  213 . 
   The first local oscillator  211  is a variable frequency oscillator with a frequency that is controlled by the control unit  105 . The second local oscillator  212  and third local oscillator  213  are fixed frequency oscillators. The frequencies F 1  of the first local oscillator  211 , and F 2  of the second local oscillator  212 , are substantially higher than the frequency F 3  of the third local oscillator  213 . 
   In operation, the input signal of the spectrum analyzer of  FIG. 2  is mixed with the first local signal  208  by the first mixer  200 , so that signals having both sum and difference frequencies of the first local signal  208  and the input signal are produced. The first band pass filter  201  selects the difference signal creating the first intermediate frequency (IF) signal  205 . 
   The first IF signal  205  is provided to the second mixer  202 , where it is mixed with the second local signal  209 . The second mixer  202  produces signals having both sum and difference frequencies of the first IF signal  205  and second local signal  209 . The second band pass filter  203  selects the difference signal creating the second IF signal  206 . 
   Similarly, the second IF signal  206  is provided to the third mixer  204  where it is mixed with the third local signal  210 . The third mixer  204  produces signals having both sum and difference frequencies of the second IF signal  206  and third local signal  210 . Low pass filter  101  selects the difference signal, creating ADC input signal  207 . 
   At each next step of spectrum measurement with the start frequency Fst, control unit  105  sets the frequency F 1  of the first local oscillator to equal F 1 =Fst+F 2 +F 3 . If the input signal has a frequency Fin, then the first IF signal  205  has a frequency F 1 −Fin, the second IF signal  206  has a frequency F 1 −Fin−F 2  and the ADC input signal  207  has a frequency F 3 −(F 1 −Fin−F 2 )=F 3 −F 1 +Fin+F 2 =F 3 −(Fst+F 2 +F 3 )+Fin+F 2 =Fin−Fst. Thus, the frequency band Fst . . . Fst+ΔF of the input signal is transferred by the frequency converter  100  to the frequency band  0  . . . ΔF at the ADC input. 
   The frequency converter  100 , shown in  FIG. 2 , carries out the necessary frequency transfer without producing harmful spurious components. However, in order to provide the high sensitivity and resolution for the spectrum analyzer that are needed to achieve a desired measurement accuracy, the frequency converter should possess one more quality: any phase noise that is introduced in the processed signal has to be correspondingly small. 
   The phase noise manifests itself as unwanted random fluctuations in a relative phase of a signal. The phase noise originates in the local oscillators of the frequency converter and finds its way into processed signal during the mixing operations. The phase noise level of a local oscillator grows when the oscillator frequency is relatively high. Therefore, the main sources of the phase noise in the block diagram of  FIG. 2  are the first local oscillator  211  (especially when it includes either a yttrium-iron-garnet (YIG) transistor or a gallium-arsenide field effect transistor (GaAs FET) oscillator, as often is the case) and the second local oscillator  212 . The third local oscillator  213  is usually a crystal oscillator with high frequency stability and very low level of phase noise. The phase noise of the first local oscillator  211  is θ 1 (t), the phase noise of the second local oscillator  212  is θ 2 (t), and the input signal and the third local oscillator are substantially free of phase noise. Then the phase noise of the first IF signal  205  is θ 1 (t), whereas phase noise of the second IF signal  206  and phase noise of the signal  207  at the ADC input is θ 1 (t)−θ 2 (t). 
   In the prior art, different methods of phase noise suppression are used in communication receivers, measuring devices and so on. One efficient approach consists of impressing the phase noise of a noisy oscillator onto a clean oscillator. Then during the mixing operations, phase noise of the first oscillator is added and phase noise of the second oscillator is subtracted from the processed signal phase. As a result, the output signal is free of the phase noise developed in the first oscillator. Such an approach was employed, for example, in U.S. Pat. No. 4,918,748, U.S. Pat. No. 6,313,619 and U.S. Pat. No. 6,600,906. The block diagram described in U.S. Pat. No. 6,600,906 is shown in  FIG. 3 . In this patent the second local oscillator  210  is supposed to have high level of phase noise. The first local oscillator  209  is taken as having a lower frequency and a small phase noise. The passage of signals in the  FIG. 3  is basically the same as in first two stages of frequency converter  100  in the spectrum analyzer of  FIG. 2 . The distinction is that first local signal  208  is produced in  FIG. 3  not by an independent local oscillator  211 , but by mixing signals from the first local oscillator  211  and the second local oscillator  212  in the mixer  301  with the subsequent selection of the sum component by BPF  300 . Thanks to such device structure the phase noise in the first  208  and the second  209  local signals are essentially the same. In the mixer  200  the phase noise of the first local signal is added to the processed signal and in the mixer  202  the phase noise of the second local signal is subtracted from the processed signal. Thus, in the mixer  202  a mutual cancellation of the phase noise of the IF signal and the phase noise of second local signal occurs. The resulting output signal has a small level of residual phase noise. In an example presented in said patent, the frequency of the input signal lies in the range from 10 MHz to 2.9 GHz, the frequency of the first local oscillator  209  varies from 505 MHz to 3.395 GHz and the frequency of the second local oscillator  210  equals 3.6 GHz. The frequency of the signal at the output of the BPF  300  equals the sum of the frequencies of the first local oscillator  209  and the second local oscillator  210 . When the frequency of the first local oscillator  209  varies from 505 MHz to 3.395 GHz, the frequency of the signal at the output of the BPF  300  is changed from 4.105 GHz to 6.995 GHz. The BPF  300  should pass all frequencies from the mentioned range and suppress the frequencies bellow 4.105 GHz. BPF  201  passes frequencies in the neighborhood of 4.095 GHz. The output signal has a frequency 495 MHz. 
   The most important reason that prevents the use of the outlined method of the phase noise suppression in a spectrum analyzer, is the appearance of numerous spurious components in the processed signal. In the context of previous example let us suppose that the frequency of the first local oscillator  211  is set up equal to 3.0 GHz (see  FIG. 4 ). The frequency of the second local oscillator  212  is fixed and equal to 3.6 GHz. After mixing in mixer  301  and selection in BPF  300 , the true first local signal  208  is created with the frequency 3.0 GHz+3.6 GHz=6.6 GHz. However, due to inevitable non-linearity in the mixer  301 , a second harmonic of the first local oscillator signal with the frequency 6.0 GHz appears at the output of the mixer  301  as well. After passing through BPF  300 , it appears as a false component of the first local signal  208  at the input of the mixer  200  ( FIG. 4   c ). Since the passband of the BPF  300  inevitably embraces the range 4.105–6.995 GHz, the true component 6.6 GHz cannot be separated from the false one 6.0 GHz by filtering. Let the input signal of the spectrum analyzer have frequency components of 1.9 GHz and 2.505 GHz. The frequency component 2.505 GHz passes the first mixer  200  and the BPF  201 , appearing in the first intermediate signal  205  as a component with a frequency 6.6 GHz−2.505 GHz=4.095 GHz. The frequency component 1.9 GHz interacts in the mixer  200  with the second harmonic 6.0 GHz and causes the appearance of the component with the frequency 6.0 GHz−1.9 GHz=4.1 GHz ( FIG. 4   e ). After frequency conversion in the mixer  202  and BPF  203  a true component 0.495 GHz and a false component 0.5 GHz are produced ( FIG. 4   f ). In this way, a by-product satellite that is unavailable in the input signal of the spectrum analyzer appears near the true component. The results of the spectrum measurements become contrary to fact and that cannot be tolerated. 
   As evidenced by forgoing discussion, a spectrum analyzer that carries out suppression of the phase noise of the local oscillators and, at the same time, does not create spurious responses in the processed signal would be an significant improvement in the art. 
   SUMMARY OF THE INVENTION 
   It is an object of the present invention to provide a spectrum analyzer with suppression of the phase noise of the local oscillators and without creating any spurious responses. 
   It is another object of the present invention to provide a spectrum analyzer with suppression of the phase noise of the local oscillators and without too stringent requirement to the filter selectivity factor that may make difficult or impossible the filters manufacturing. 
   It is a further object of the present invention to provide a spectrum analyzer where incomplete compensation of the phase noise is precluded. 
   In order to accomplish the first object of the present invention, a phase noise compensation unit is incorporated in a spectrum analyzer having a three stage frequency converter at its input. Two signal inputs of the phase noise compensation unit are connected to the outputs of the first local oscillator and the second local oscillator. The control input of the phase noise compensation unit is connected to the output of the control unit of the spectrum analyzer. The output of the phase noise compensation unit is connected to the input of the third mixer of the frequency converter. The phase noise compensation unit processes the first and the second local signals that are applied to its signal inputs with the use of the information about current start frequency which is received through the control input. The phase noise compensation unit produces at its output a signal that has a frequency equal to the desired frequency of the third local signal. The phase noise of this signal is made equal to the difference θ 1 (t)−θ 2 (t), where θ 1 (t) is the phase noise of the first local oscillator of the frequency converter and θ 2 (t) is the phase noise of the second local oscillator. At the same time the phase noise compensation unit ensures that its output signal is free from any spurious responses. 
   The second IF signal of the frequency converter contains the same phase noise θ 1 (t)−θ 2 (t). The third mixer of the frequency converter mixes the second IF signal and the third local signal. During this process phase noise of the second IF signal and phase noise, inserted in the third local signal, cancel each other. Therefore, the resulting signal that comes to the input of the ADC, has a negligibly small phase noise. 
   The first and the second local signals are applied to the first and the second mixers from the first and the second local oscillators directly, so that they do not have spurious responses. The third local signal does not have spurious responses thanks to the precautions, which are taken in the phase noise compensation unit. Hence, the resulting signal that comes to the input of the ADC, is free of any spurious responses as well. 
   According to the present invention the phase noise compensation unit consists of a first auxiliary frequency converter, a second auxiliary frequency converter and a reference oscillator. The first auxiliary frequency converter processes the first and the second local signals, creating a signal that has a phase noise θ 1 (t)−θ 2 (t). The reference oscillator at each step of spectrum measurement generates under control from the control unit a signal with a frequency that equals the start frequency. The second auxiliary frequency converter uses the output signals of the first auxiliary frequency converter and the reference oscillator to produce the third local signal with the desired frequency and with the same phase noise θ 1 (t)−θ 2 (t). Simultaneously, the second auxiliary frequency converter removes from the output signal all spurious components that could appear during the mixing operations. 
   According to the present invention, each of the two auxiliary frequency converters includes a mixer in series with a filter. Such an assembly produces an output signal that has a frequency, equal to the difference of the frequencies of the two input signals. 
   In order to accomplish the second object of the present invention, the phase noise compensation unit is complemented by a third auxiliary frequency converter that has an input and an output. The input of the third auxiliary frequency converter is connected to the output of the phase noise compensation unit. The output of the third auxiliary frequency converter is connected to the input of the third mixer of the frequency converter. The third auxiliary frequency converter comprises a mixer, a band pass filter and a third local oscillator. It creates at its output, a signal with a frequency that equals the sum of the frequency of the input signal and the frequency of the third local oscillator. The first input of the mixer is used as the input of the third auxiliary frequency converter, the second input of the mixer is connected to the output of the third local oscillator. The output of the mixer is connected to the input of the band pass filter. The output of the band pass filter is used as the output of the third auxiliary frequency converter. 
   In order to accomplish the third object of the present invention, a delay line is inserted in the path of the processed signal in the frequency converter before the third stage of conversion. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a block diagram of a prior art spectrum analyzer. 
       FIG. 2  is a block diagram of a prior art spectrum analyzer with a three-stage frequency converter. 
       FIG. 3  is a block diagram of a prior art frequency converter with phase noise compensation. 
       FIG. 4  illustrates the appearance of spurious responses in a prior art frequency converter with phase noise compensation. 
       FIG. 5  is a block diagram of a spectrum analyzer according to the present invention. 
       FIG. 6  is a block diagram of a spectrum analyzer according to the present invention, with the inner structure of the phase noise compensation unit being disclosed. 
       FIG. 7  illustrates the suppression of the spurious responses in a spectrum analyzer according to the present invention. 
       FIG. 8  is a block diagram of a spectrum analyzer according to another embodiment of the present invention. 
       FIG. 9  is a block diagram of a spectrum analyzer according to yet another embodiment of the present invention. 
   

   DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
   A block diagram of a basic embodiment of a spectrum analyzer according to the present invention is shown at  FIG. 5 . In  FIG. 5 , the frequency converter  100 ′ differs from that in  FIG. 1 , but the remaining blocks  101 ,  102 ,  103 ,  104 , and  105  can be similar to the correspondingly numbered blocks in  FIG. 1 . The frequency converter  100 ′ includes a first mixer  200 , a first band pass filter (BPF)  201 , a second mixer  202 , a second band pass filter (BPF)  203 , a third mixer  204 , the first local (variable frequency) oscillator  211  and the second (fixed frequency) local oscillator  212 , all similar to correspondingly numbered elements in  FIG. 2 . 
   A phase noise compensation unit  500  is incorporated in the frequency converter  101 ′ of spectrum analyzer. Two signal inputs  501  and  502  of the phase noise compensation unit  500  are connected to the outputs of the first local oscillator  211  and the second local oscillator  212  respectfully. A control input  503  of the phase noise compensation unit  500  is connected to the output of the control unit  105 . The output  504  of the phase noise compensation unit  500  is connected to the input of the third mixer  204 . The phase noise compensation unit  500  processes the first local signal  208  and the second local signal  209  with the use of the information about current start frequency Fst received through the control input  503 . As a result, an output signal  504  is produced, this signal having a frequency equal to the desired frequency of the third local signal  210 . The phase noise compensation unit  500  inserts in the output signal  504  a phase noise that equals the difference θ 1 (t)−θ 2 (t) between the phase noise θ 1 (t) of the first local oscillator  211  and the phase noise θ 2 (t) of the second local oscillator  212 . What is most important, the phase noise compensation unit  500  produces the signal  504  free of any spurious components 
   In the first mixer  200  and the first BPF  201 , the frequency of the input signal is subtracted from the frequency of the first local signal  208 . Therefore, the phase noise of the first intermediate signal  205  is the same as phase noise θ 1 (t) of the first local oscillator  211 . In the second mixer  202  and the second BPF  203 , the frequency of the second local signal  209  is subtracted from the frequency of the first intermediate signal  205 . This being so, the phase noise of the second intermediate signal  206  equals the difference θ 1 (t)−θ 2 (t) between the phase noise θ 1 (t) of the first local oscillator and the phase noise θ 2 (t) of the second local oscillator. Thus, the phase noise of the second intermediate signal  206  and the phase noise inserted by the phase noise compensation unit  500  in the third local signal  210  are the same. In the third mixer  204  and LPF  101  the phase noise of the third local signal  210  is subtracted from the phase noise of the second intermediate signal  206 . As a result a mutual cancellation of the phase noises takes place, so that the signal  207  at the input of the ADC  102  has a negligibly small phase noise. 
   In the block diagram of  FIG. 5 , the first local signal  208  and the second local signal  209  are directly applied to the respective first and the second mixers; consequently they are free from spurious responses. The third local signal  210  is free from spurious responses thanks to the precautions that are taken in the phase noise compensation unit  500 . Therefore, the resulting signal that comes to the ADC input  207  is free of any spurious responses. 
     FIG. 6  shows a block diagram of the spectrum analyzer according to the present invention, with the inner structure of the phase noise compensation unit  500  being disclosed in detail. The phase noise compensation unit  500  consists of a fourth mixer  600 , a second LPF  601 , a fifth mixer  603 , a third BPF  604  and a reference (variable frequency) oscillator RO  605 . The fourth mixer  600  and the second LPF  601  form a first auxiliary frequency converter. The fifth mixer  603  the third BPF  604  act as a second auxiliary frequency converter. 
   The first auxiliary frequency converter receives at its inputs  501  and  502  the first local signal  208  and the second local signal  209 . The output signal  602  of the first auxiliary frequency converter has a frequency that equals the difference F 1 −F 2  between frequencies of the first local signals  208  and the second local signal  209 . Accordingly, the phase noise of the signal  602  equals the difference θ 1 (t)−θ 2 (t) between the phase noise θ 1 (t) of the first local signal  208  and the phase noise θ 2 (t) of the second local signal  209 . Along with the signal  602 , the first auxiliary frequency converter produces numerous spurious components. 
   The reference oscillator  605  is a variable frequency oscillator. At each step of spectrum measurement the control unit  105  sets the frequency Fref of the reference oscillator  605  to equal the start frequency Fst of the frequency band that is analyzed at the current step. The frequency of the reference oscillator is lower than the frequencies of the first and the second local oscillators, and no limitations are imposed on the presence of spurious responses in its output signal, so that its phase noise is sufficiently small. 
   The inputs of the second auxiliary frequency converter are connected to the output  602  of the first auxiliary frequency converter and to the output  606  of the reference oscillator  605 . The frequency of the output signal  504  of the second auxiliary frequency converter equals the difference between the frequency of the signal  602  and the frequency of the reference oscillator  605  and equals (F 1 −F 2 )−Fref=((Fst+F 2 +F 3 )−F 2 )−Fst=F 3 . Thus, the frequency of the signal  504  at the output of the second auxiliary frequency converter or, what is the same, at the output of phase noise compensation unit  500  equals the desired frequency of the third local signal. Since the output signal of the reference oscillator is free of phase noise, the phase noise of the signal  504  equals the phase noise of the signal  602 . Therefore, the phase noise of the signal  504  equals the difference θ 1 (t)−θ 2 (t) between the phase noise θ 1 (t) of the first local signal  208  and the phase noise θ 2 (t) of the second local signal  209 . An important function of the second auxiliary frequency converter is the clearing the output signal  504  from all spurious components. 
   The third BPF  604  has a bandwidth that is equal to or less than a common divisor F 0  of the local oscillators frequencies F 1 , F 2  and the reference oscillator frequency Fref. The frequencies of the spurious components that emerge in the mixers  600  and  603  constitute linear combinations of the frequencies of these mixers input signals. Since the frequencies of the local oscillators and the reference oscillator are multiples of the frequency F 0 , the frequencies of the mentioned spurious components are multiples of the frequency F 0  as well. The situation is illustrated in  FIG. 7 . The distance between a spurious component and the signal  504  in the frequency domain is k*F 0 , where k is an integer not less than 1. This distance cannot be less than F 0 . On the other hand, the third BPF  604  suppresses all components that are farther than F 0 /2 from the frequency of the signal  504  and are out of the filter pass band. As a result, the third BPF  604  allows passage of the signal  504  and suppresses all spurious responses, so that the signal  504  is free of spurious responses completely. 
   The present invention may be best understood by way of a specific example. In this example the frequency range of the input signal of the spectrum analyzer is 0 . . . 3000 MHz. The sampling rate of the ADC  102  is 100 MHz. The cutoff frequency of the LPF  101  is accordingly 35 MHz and the spacing between adjacent start frequencies is ΔF=25 MHz. When the spectrum of the input signal is measured step by step, the start frequency Fst takes on values 0, 25 MHz, 50 MHz, . . . , k*25 MHz, . . . , 2975 MHz. The frequency F 1  of the first local oscillator is set accordingly as 6500 MHz, 6525 MHz, . . . , 9475 MHz. The frequencies F 2  and F 3  of the second and the third local signals are fixed and equal 5500 MHz and 1000 MHz respectfully. At each step of the spectrum measurement the relationship F 1 =Fst+F 2 +F 3  is held. The frequency of the signal  602  at the output of the second LPF  601  takes on values F 1 −F 2 =1000 MHz, 1025 MHz, . . . , 3975 MHz. The frequency of the reference oscillator  605  is set at each step by the control unit to be equal to 0, 25 MHz, 50 MHz, . . . , 2975 MHz. The frequency of the signal  504  at the output of the third BPF  604  equals the difference between frequencies of the signal  602  and the reference oscillator  605 ; this frequency remains fixed at the value 1000 MHz. The third BPF  604  represents a filter with a central frequency 1000 MHz and a bandwidth 25 MHz. Such a filter allows passage of frequencies from 987.5 MHz up to 1012.5 MHz and suppresses all frequencies that are out of this band. It is easy to see that frequencies of all signals in the spectrum analyzer of the cited example are multiples of 25 MHz. For this reason, the frequencies of all spurious components that appear in the mixers  600  and  603  are multiples of 25 MHz as well. The spurious component, which is the closest to the central frequency 1000 MHz of the third BPF  604 , may have a frequency 975 MHz or 1025 MHz. But these frequencies lie outside the pass band of the third BPF  604 , therefore they (as well as all other spurious components) are suppressed by this filter. 
   It may happen that the common divisor F 0  of the local oscillators frequencies F 1 , F 2  and the reference oscillator frequency Fref is relatively small. The value of the common divisor F 0  dictates the bandwidth of the BPF  604 . When the common divisor F 0  and, accordingly, the bandwidth of the BPF  604  are too small, the required filter selectivity factor increases, and it becomes difficult or impossible to manufacture needed filter. 
     FIG. 8  shows a block diagram of another embodiment of the present invention. This block diagram includes a frequency converter  110 ″, the purpose of which is to overcome the mentioned difficulty. Here, a third auxiliary frequency converter, that consists of a sixth mixer  801 , a forth BPF  800  and the third local oscillator  213 , is inserted between the output  504  of the phase noise compensation unit  500  and the input of the third mixer  204 . The output  504  of the third BPF  604  and the output of the third local oscillator  213  are connected to the inputs of the mixer  801 . The mixer  801  creates at its output signals having both sum and difference frequencies of the input signals. The forth BPF  800  selects the sum product and passes it to the input of the third mixer  204  as the third local signal  210 . In this embodiment of the present invention at each measurement step, the control unit  105  sets the frequency Fref of the reference oscillator  605  to be equal to: Fref=(F 1 −F 2 )−Foff=Fst+F 3 −Foff. Here, F 3  is the desired frequency of the third local signal  210  and Foff is an offset frequency. The offset frequency Foff is chosen as a divisor of the frequency F 3 . The frequency of the third local oscillator  213  is made equal to F 3 −Foff. The bandwidth of the forth BPF  800  is equal to or less than the offset frequency Foff. 
   The frequency of the output signal  504  of the third BPF  604  equals the difference between the frequency of the signal  602  (that equals F 1 −F 2 ) and the frequency Fref of the reference oscillator  605 . Taking in account the relationships F 1 =Fst+F 2 +F 3  and Fref=Fst+F 3 −Foff, it is easy to see, that the frequency of the output signal  504  of the third BPF  604  equals (F 1 −F 2 )−Fref=((Fst+F 2 +F 3 )−F 2 )−(Fst+F 3 −Foff)=Foff. Therefore, the third BPF  604  has a central frequency Foff and bandwidth F 0 ; its filter selectivity factor equals Foff/F 0 . By choosing the proper value of the offset frequency Foff the ratio Foff/F 0  may be reduced in an arbitrary way, so that the manufacturing of the third BPF  604  does not present any difficulties. As before, the output signal  504  of the third BPF  604  has a phase noise θ 1 (t)−θ 2 (t) and is free of any spurious components 
   During mixing process in the sixth mixer  801 , some new spurious components emerge. The frequencies of these spurious components constitute linear combinations of the frequency Foff of the signal  504  and the frequency F 3 −Foff of the third local oscillator  213 . Since the frequencies F 3  and F 3 −Foff are multiples of the frequency Foff, the frequencies of the spurious components appearing in the sixth mixer  901  are multiples of the frequency Foff as well. The distance between a spurious component and the signal  210  is k* Foff, where k is an integer not less than 1. This distance cannot be less than Foff. On the other hand, the bandwidth of the forth BPF  800  is equal to or less than the offset frequency Foff. Accordingly, the forth BPF  800  suppresses all components that are farther than Foff/2 from the signal  210  and are out of the filter pass band. As a result, the forth BPF  800  allows passage of the signal  210  and suppresses all spurious responses that appeared in the sixth mixer  801 . 
   The frequency of the third local oscillator  213  is much less than the frequencies of the first and the second local oscillators, therefore it has essentially zero phase noise. For this reason, the phase noise in the third local signal  210  is the same as in the signal  504  and equals θ 1 (t)−θ 2 (t). 
   The frequency of the signal  210  equals the sum of the frequency Foff of the signal  504  and the frequency F 3 −Foff of the third local oscillator  213  and equals Foff+(F 3 −Foff)=F 3 . Besides, as it was just mentioned, the signal  210  has a phase noise θ 1 (t)−θ 2 (t) and is free of spurious responses completely. Thus, the block diagram of  FIG. 8  furnishes all necessary features of the third local signal  210  alleviating at the same time the requirements to the third BPF  604 . 
     FIG. 9  shows a block diagram of yet another embodiment of the present invention, including a frequency converter  100 ′″, the purpose of which is to eliminate a possibility of incomplete phase noise compensation. The bandwidth of the third BPF  604  may be narrower than bandwidth of the second BPF  203 . Because of it, the time delay of BPF  604  may exceed considerably the time delay of BPF  203 . The same phase noise comes to the third mixer  204  through two routes: through BPF  203  and through BPF  604 . If the delays in these two routes are different, then the mutual cancellation of the phase noises in the third mixer  204  is not complete, and residual phase noise penetrates into the signal  207  at ADC input. To prevent an appearance of such residual phase noise a proper delay line  902  is inserted between the output of the second BPF  203  and the input of the third mixer  204 . 
   A number of implementations of the present invention were described above. It should be apparent to those skilled in the art that various modifications are possible without departing from the principles of the present invention. Accordingly, such modifications are understood to be within the scope of the following claims.