Abstract:
A drive circuit for an alternate-phase brushless DC motor is arranged to minimize the torque ripple normally occurring with such motors. The motor has rotor magnets and at least two stator coils, and has a torque relationship in which the rotational torque for each coil extends over an electrical angle of more than 180 degrees and has gradual rising and falling portions intersecting with respective falling and rising portions of the torque of the other coil. The drive circuit includes a rotary position sensor, such as a Hall element, switching transistors for driving the stator coils, and switching signal forming circuitry responsive to the sensor to switch the switching transistors on and off alternately. In order to reduce the torque ripple, the drive circuit is made to energize the stator coils in the region of the intersections of the rotational torque, so that the coils are energized for more than 180 electrical degrees. In one preferred embodiment, a pair of additional switching transistors coupled to the stator coils are switched on by a circuit formed of a pair of differential amplifiers coupled to the Hall element and, through a logic gate, to the additional switching transistors.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     This invention is directed to a drive circuit for an alternate phase brushless DC motor, and more particularly relates to a drive circuit arranged to minimize torque ripple. 
     2. Brief Description of the Prior Art 
     It has become common to reduce wow and flutter in VTRs cassette tape recorders, and the like, by incorporating therein a &#34;direct drive system&#34; including a superslow speed brushless DC motor directly connected to a reel mount for directly driving the latter. The so-called &#34;direct drive system&#34; also makes it possible to improve the operability and reliability of the system and to reduce the size of the system. However, a brushless motor having a remarkably small torque ripple has to be used to ensure that the wow and flutter are reduced. For this purpose, a brushless motor having an exceptionally high performance quality is necessary, but the narrow tolerances required to achieve such performance quality for that motor raise the production cost of the brushless DC motor to an unrealistically high level, and thereby make it difficult to put such a motor into practical use. 
     Therefore, with a view to reducing torque ripple, previous proposals have been made, including using non-conventional techniques for winding the armature coils, forming armature coils into non-conventional shapes, and altering the magnetizing pattern of the rotor magnets. However, while these proposals have had some success in reducing the torque ripple, such success has been limited and insufficient. 
     The following types of DC brushless motors are well known, and each has its own well known intrinsic operating characteristics and its own peculiar drawbacks: 
     (1) Three-Phase Unidirectional Energizing Type 
     In this type motor, three-phase coils are successively supplied with current in one direction. The motor of this type has a relatively simple circuit construction but has a drawback in that the torque ripple is quite high. 
     (2) Three-Phase Bidirectional Energizing Type 
     In this type motor, a selected two out of three three-phase coils are consecutively supplied with current in a selected direction, with the third coil being supplied with current in a constant direction, thereby to effect bidirectional operation. The motor of this type has its torque ripple reduced more than that of the type (1) but has a drawback in that its drive circuit construction is rather complex. 
     In motors of either of the aforementioned types (1) and (2), since the current change-over timing and the pole inverting timing overlap with each other, they each have a common drawback in that the torque drops upon current changeover. 
     (3) Sine-Wave Drive Type 
     In this type motor, a sinusoidal drive current, having a strength that varies in proportion to the linking magnetic fluxes of the respective alternate coils, is supplied to these coils so that the sum of the torques generated by the drive currents and the linking magnetic fluxes is a constant independent of the rotational angle of the rotor. The motor of this type can suppress the torque ripple to a rather low level, but requires extremely accurate adjustment of all circuit parameters. Furthermore, because it requires a double power source, which cannot easily be constructed to be efficient and small in size, this motor type is unsuitable for use where a small-sized motor is demanded. 
     (4) Alternate-Phase Switching Type 
     In this type motor, a unidirectional torque is generated in each phase of first and second alternate stator coils over an angle of more than 180 electrical degrees, and the first and second stator coils are alternately switched and supplied with power each 180 electrical degrees, thereby generating a continuous rotational torque. A constant, continuous rotational torque is provided except in the neighborhood of the current change-over times of the first and second stator coils. 
     However, since in such an alternate-phase brushless DC motor the torque drops somewhat upon a current change-over (or switching) operation, as shown in FIG. 2A below, it has a drawback in that unacceptable torque ripple results. Moreover, if this type of brushless DC motor is used, for example, as the capstan driving motor of a VTR, the wow and flutter that can result from this torque ripple, which has a significant high-frequency component, can result in jitter. 
     OBJECTS AND SUMMARY OF THE INVENTION 
     It is therefore an object of the present invention to provide a drive circuit for an alternate phase brushless DC motor so that the above-mentioned torque drop, which normally occurs upon the current change-over, is reduced. 
     Another object of the present invention is to reduce wow and flutter in a VTR, tape recorder, or the like, and to reduce the incidence of jitter, by providing therein an alternate-phase brushless DC motor incorporating the drive circuit of the present invention. 
     In accordance with an aspect of this invention, for each of the first and second stator coils, an overlap energizing current is provided in the neighborhood of the times of current change-over, to compensate for the torque drop that normally occurs at such times. 
     More specifically, in an embodiment of this invention, a drive circuit for an alternate phase brushless DC motor having at least first and second stator coils of alternate phase and a rotor including rotor magnets, the stator coils and the rotor magnets enjoying a rotational torque relationship such that the rotational torque for each coil extends over an electrical angle of more than 180 degrees and has gradual rising and falling portions having points of intersection with falling and rising portions, respectively, of the rotational torque of the other of the coils, comprises a sensing device, such as a Hall element, to detect the rotational position of the rotor magnets, a switching signal circuit providing alternate switching signals in response to the rotational position detected by the sensing device, and a pair of switching devices respectively coupled to the stator coils to alternately energize the latter in response to the respective alternate switching signals applied thereto. In order to reduce the torque ripple, the drive circuit includes torque correcting circuitry to cause an overlap energizing current flow through the first and second stator coils during the times that the rotor is in the vicinity of the above-mentioned points of intersection, so that the energizations of the stator coils overlap each other in phase. This torque correcting circuitry can include an additional pair of switching devices coupled to the respective stator coils for energizing the same prior to and following such points of intersection. 
     Other objects, features, and advantages of the invention will be apparent from the following description when read in conjunction with the accompanying drawings. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a cross-sectional view showing a four-pole alternate phase DC brushless motor to which this invention can be applied; 
     FIG. 2A and FIG. 2B are waveform diagrams illustrating the rotational torque pattern of the motor of FIG. 1; 
     FIG. 3 is a circuit diagram showing a motor driving circuit according to a first embodiment of this invention; 
     FIG. 4A to FIG. 4F are waveform diagrams for explaining the operation of the motor driving circuit of FIG. 3; 
     FIG. 5A to FIG. 5C are waveform diagrams for explaining the operation of the motor driving circuit of FIG. 3; 
     FIG. 6A and FIG. 6B are circuit diagrams showing a motor driving circuit according to a second embodiment of this invention, in which the present invention is applied to the supply reel motor of a VTR; 
     FIG. 7A to FIG. 7E are waveform diagrams for illustrating the operation of the motor driving circuit of FIG. 6A and FIG. 6B; 
     FIG. 8 is a circuit diagram showing a motor driving circuit according to a third embodiment of this invention; and 
     FIG. 9A to FIG. 9E are diagrams for illustrating the operation of the motor driving circuit of FIG. 8. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     FIG. 1 is a cross-sectional view showing one example of an alternate phase brushless DC motor to which this invention can be applied. In this motor, a cylindrical outer yoke 1 has four permanent rotor magnets 2a, 2b, 3a and 3b fixed on its inner circumferential surface, while a stator core 4 is arranged coaxially with the rotor yoke 1 and has stator coils 5A and 5B, and 5A&#39; and 5B&#39; attached to its circumferential surface. Further, the stator coils 5A and 5A&#39; are connected in series and are of the same electrical phase (hereinafter, phase A), while, similarly, the stator coils 5B and 5B&#39; are also connected in series with each other and are of the same electrical phase (phase B). 
     The rotor magnets 2a, 2b, 3a and 3b each subtend an electrical angle of 140°. For example, if a boundary between the north polar magnets N and south polar magnets S is defined as an angular position or rotational angle of 0°, air gaps 6a and 6b, each occupying angles electrically extending from 140° to 220° are formed opposite to each other. Respective backward and forward current path portions U and V of the coils 5A and 5B, and 5A&#39; and 5B&#39; are separated from each other by an electrical angle of 100°. 
     It will be understood that the air gaps 6a and 6b serve to reduce the linking magnetic flux in the vicinity of an electrical angle of 180°. 
     FIG. 2A is a plot of the rotational torque generated by the A-phase stator coils (i.e., coils 5A and 5A&#39;). Magnetic flux linking the forward current path portion V of the A-phase coils 5A and 5A&#39; changes as shown by a thin line curve A&#39;. 
     If current is continuously applied through the A-phase stator coils 5A and 5A&#39; (i.e., for an electrical angle of 720°), a rotational torque shown by the curve A&#39; is attributable to the forward current path portion V. At the same time, the current flowing through the backward current path portion U of the A-phase coils 5A and 5A&#39; is opposite to the current flowing through the forward current path portion V thereof, and differs in phase therefrom by an electrical angle of 100°. Accordingly, rotational torque as shown by a dotted line curve B&#39; is attributable to the backward current path portion U. A composite rotational torque shown by a solid line curve A is generated by the combined forward and backward current path portions V and U of the A-phase coils 5A and 5A&#39;. In a conventional alternate phase DC brushless motor the stator coils 5A and 5A&#39; are energized over an electrical angle of 180° corresponding to the span illustrated by the vertical chain lines on FIG. 2A, although the composite rotational torque A has the same sense or direction over an electrical angle exceeding 180°. 
     It will be observed that the rotational torque A is substantially flat over an electrical angle of 100°, and has gradually rising and falling portions extending the rotational torque to and beyond the above span of 180° during which current is normally applied to the stator coils. 
     The B-phase stator coils (i.e., stator coils 5B and 5B&#39;) produce a similar rotational torque, but offset by an electrical angle of 180° from that of coils 5A and 5A&#39;. The rotational torque attributable to coils 5B and 5B&#39; has a flat portion and gradually rising and falling portions having points of intersection with the falling and rising portions, respectively, of the rotational torque A attributable to stator coils 5A and 5A&#39;. 
     The stator coils 5B and 5B&#39; are also conventionally energized over spans of 180° electrical angle alternately with the stator coils 5A and 5A&#39;. Normally, the current change-over between the A-phase stator coils 5A and 5A&#39; and the B-phase stator coils 5B and 5B&#39; occurs at the above-mentioned points of intersection of the rotational torque. 
     The total composite rotational torque from all the stator coils 5A, 5A&#39;, 5B, and 5B&#39; thus has the relation depicted in FIG. 2B. 
     However, since the torque drops in the neighborhood the current changeover time (i.e., when switched), as shown in FIG. 2B, the conventional brushless DC motor thus far described has a drawback in that torque ripple occurs. Moreover, if this brushless motor is used as the capstan driving motor of a VTR, for example, the wow and flutter caused by the torque ripple have a significant high frequency component, which can cause jitter. 
     This invention eliminates the aforementioned disadvantage concomitant with the alternate phase switching type brushless DC motor and provides an alternate phase motor driving circuit that successively reduces the drop in torque occurring when the stator current is changed over by overlapping the respective energizing periods of the A-phase and B-phase stator coils 5A, 5A&#39;, and 5B, 5B&#39;. 
     Several perferred embodiments of this invention will now be described with reference to FIG. 3 to FIG. 9. 
     FIG. 3 shows a motor driving circuit according to a first embodiment of this invention. Although not shown in FIG. 1, the brushless DC motor is equipped with position detecting magnets disposed on its rotor and with Hall elements or other equivalent detectors each opposite thereto and positioned on its stator. 
     In the drive circuit shown in FIG. 3, a Hall element 10 generates alternate output signals A 1  and B 1  (FIG. 4A) at outputs A and B thereof and such signals are applied to respective inputs of a differential amplifier 11. A current changeover signal a (FIG. 4B) is provided from an uninverted output terminal Q of the differential amplifier 11 while a complementary current change-over signal b (FIG. 4C) is provided from an inverted output terminal Q thereof. The change-over signals a and b alternate between high and low levels at intervals corresponding to electrical angles of 180°. 
     Current switching transistors Tr 1  and Tr 2  have their collectors respectively coupled in series with the A-phase and B-phase stator coils 5A, 5A&#39; and 5B, 5B&#39;, and have their bases respectively coupled to receive the change-over signals a and b. 
     The stator coils 5A, 5A&#39; and 5B, 5B&#39; are also coupled to a power source +B. 
     Thus the transistors Tr 1  and Tr 2  are alternately turned on and off by the aforementioned signals a and b, so that the A-phase stator coils 5A and 5A&#39; and the B-phase stator coils 5B and 5B&#39; are intermittently energized, with each period of energization corresponding to an electrical angle of 180°. Consequently, the motor is driven in a predetermined rotational direction. 
     Meanwhile, the AC output signals A 1  and B 1 , as shown in FIG. 4A, are furnished from the Hall element 10 to inputs of each of a pair of differential amplifiers 13 and 14. These differential amplifiers 13 and 14 are designed to have a predetermined threshold level as indicated by chain line in FIG. 4A. Accordingly, from the one differential amplifier 13 there is generated an output signal c (FIG. 4D), which takes on a high level when the level of the output signal A 1  is higher than the threshold level. Likewise, a corresponding output signal d (FIG. 4E) is generated from the other differential amplifier 14. Those output signals c and d are fed to inputs of an AND circuit 15, which in turn provides a pulse signal e (FIG. 4F) which takes on a high level in the vicinity of the points of intersection of the rotational torques, i.e., slightly before and slightly after the edges of the aforementioned current change-over signals a and b. The pulse signal e from the AND circuit 15 is fed to each of the bases of a pair of torque-correction transistors Tr 3  and Tr 4  which have their collectors respectively coupled through collector resistors R 9  and R 8  to the stator coils 5A, 5A&#39; and 5B, 5b&#39;. 
     Thus the transistors Tr 3  and Tr 4 , which are rendered conductive when the pulse signal e is applied at a high level, serve to supply an overlap energizing current to the stator coils 5A, 5A&#39; aND 5B, 5B&#39; to compensate the above-mentioned torque drop. 
     As a result, in the motor driving circuit thus constructed, the A-phase stator coils 5A and 5A&#39; are energized over the period for which the transistor Tr 1  is kept conductive by the current change-over signal a from the differential amplifier 11, as shown in FIG. 5A. Moreover, as shown in FIG. 5B, immediately before the transistor Tr 1  is switched from its conductive to its non-conductive state in response to the aforementioned current change-over signal a, the transistor Tr 4  is rendered conductive by the pulse signal e so that the B-phase stator coils 5B and 5B&#39; are energized over a predetermined period t immediately before the time of current change-over. Correspondingly, if the transistor Tr 1  is switched to its non-conductive state whereas the transistor Tr 2  is switched to its conductive state, the B-phase stator coils 5B and 5B&#39; are energized by the current change-over signal b. At this current change-over time, since the transistor Tr 3  is rendered conductive by the pulse signal e, the A-phase stator coils 5A and 5A&#39; are energized over the predetermined period t immediately after the current is changed over, as is also shown in FIG. 5B. 
     As result, since the A-phase stator coils 5A and 5A&#39;  and the B-phase stator coils 5B and 5B&#39; are energized together in an overlapping manner in the neighborhood of (i.e., before and after) the instant when the phase shift is accomplished from A to B or from B to A, the torque of the motor has its drop compensated during current change-over. As indicated at the dotted curve line in FIG. 5C, the torque characteristics are flattened. That is, the &#34;valleys&#34; between A and B current phases in the torque curve shown in FIG. 2B are filled in due to energization of the transistors Tr 3  and Tr 4 . 
     Incidentally, the compensation of the torque drop can be set at any desired value by selecting the values of the respective collector resistors R 9  and R 8  of the transistors Tr 3  and Tr 4 . 
     It should also be recognized that although the transistors Tr 3  and Tr 4  are turned on and rendered conductive, they do not both carry current at the same time. For example, when the transistor Tr 1  is conductive, that transistor dominates and no current flows through the transistor Tr 3 . Similarly, current flows through the transistor Tr 4  only before and after the energizing period of the transistor Tr 2 . 
     FIG. 6A and FIG. 6B, and FIG. 7A to FIG. 7E are presented for explaining another embodiment, in which the present invention is applied to a reel motor driving circuit for a magnetic tape recorder. 
     Elements in this embodiment that are common to the embodiments of FIG. 3 are identified with the same reference characters, and a detailed description thereof is omitted. 
     First of all, the AC output signals A 1  and B 1 , as shown in FIG. 7A, which are generated from the output terminals A and B of the Hall element 10 in response to the detected rotation of the rotor magnets 2A, 2B, 3A and 3B, are fed to the bases of transistors Tr 5  and Tr 6  which are arranged as a differential amplifier. As a result, a pulsating current change-over signal f, as shown in FIG. 7B, is generated from the collector of the transistor Tr 6 . This current change-over signal f is fed to the base of a transistor Tr 7  whose collector is coupled to the base of another transistor Tr 8  so that the transistors Tr 7  and Tr 8  are alternately turned on and off every electrical angle of 180°. 
     A switching transistor Tr 9  has its collector coupled to the collectors of the transistors Tr 7  and Tr 8  and has a control voltage, as appropriate to the forward and backward rotation of the supply reel motor, applied from a power source terminal 20 to its base. The transistor Tr 9  is switched on and off in accordance with this control voltage. 
     When the supply reel motor is rotating in the forward direction (i.e., for a tape-rewinding operation), a voltage at a high level is applied from the power source terminal 20 to the base of the transistor Tr 9  and thereby renders the transistor Tr 9  conductive. During the backward rotations (for a fast-forward operation), a voltage at a low level is fed from the power source terminal 20 to the base of the transistor Tr 9  and thereby switches the transistor Tr 9  to its non-conductive state. 
     Initially, the operation of the circuit for forward rotation of the motor will be described. 
     A switch SW has a movable contact coupled to the stator coils 5A, 5A&#39;, 5B and 5B&#39; and has a fixed terminal V R  coupled to the power source terminal 20 and another fixed terminal V S  connected to receive a rotational speed controlling voltage, which can be the output of a tape tension servo. 
     For forward operation the movable contact of the switch SW is coupled to the terminal V S , and a high level is applied from the power source terminal 20 to the transistor Tr 9  to render it conductive. 
     The transistors Tr 7  and Tr 8  provide, from their respective collectors, pulse signals according to the aforementioned current change-over signal f, and the respective pulse signals are fed to a current change-over circuit formed of transistors Tr 10  to Tr 15 , resistors R 1  to R 4 , and capacitors C 1  to C 3 . Specifically, the aforementioned respective pulse signals are applied to the bases of the transistors Tr 11  and Tr 12 , which have their emitters coupled to the collector of the transistor Tr 10 , and whose collectors are coupled to the bases of the switching transistors Tr 15  and Tr 14 , respectively. These switching transistors Tr 14  and Tr 15  are connected to energize the stator coils 5A, 5A&#39; and 5B, 5B&#39;, respectively. 
     The resistors R 1  and R 2  respectively connect the base and emitter of the transistor Tr 10  to a supply voltage V CC , while the resistors R 3  and R 4  are respectively coupled from ground to the bases of the transistors Tr 11  and Tr 12  to bias the same. The capacitor C 1  is coupled in parallel with the resistor R 3 . 
     If a pulse signal at a low level is fed from the collector of the transistor Tr 7  to the base of the transistor Tr 11 , this transistor Tr 11  is rendered conductive so that a current I flows through the resistor R 2  and the transistors Tr 10  and Tr 11 . As a result, the transistor Tr 15  is turned on. Correspondingly, if a pulse signal at a low level is fed from the collector of the transistor Tr 8  to the base of the transistor Tr 12 , this transistor Tr 12  is turned on to then actuate the transistor Tr 14 . 
     Thus, the A-phase coils 5A and 5A&#39; and the B-phase coils 5B and 5B&#39; are alternately energized. Incidentally, the capacitors C 2  and C 3 , here arranged in parallel with the transistors Tr 14  and Tr 15 , serve to prevent noise. 
     The collector voltages, during the time that the respective transistors Tr 14  and Tr 15  are conductive, are detected by means of a detecting circuit formed of diodes D 1 , D 2 , and D 3 , and the detected voltages are fed to a base current controlling circuit formed of the transistors Tr 13  and Tr 10  and resistors R 1  and R 2 . The diode D 1  has its cathode connected to the junction of the collector of the transistor Tr 15  with the stator coils 5B, 5B&#39; and the diode D 3  has its cathode connected to the junction of the collector of the transistor Tr 14  with the stator coils 5A, 5A&#39;. The diode D 2  has its anode joined to the anodes of the diodes D 1  and D 3  and has its cathode connected to the base of the transistor Tr 13 . 
     The detected collector voltage from the transistors Tr 14  and Tr 15  is furnished through the transistor Tr 13  to control the current flowing to the transistor Tr 10 , which, in turn, controls the amount of current flowing through the transistors Tr 11  and Tr 12 . As a result the transistors Tr 11  and Tr 12  furnish sufficient base current I B  to drive the transistors Tr 14  and Tr 15  to saturation, but do not permit such base current to become excessive. 
     The operation of this circuitry can be explained as follows. 
     The voltage at point G in FIG. 6B assumes a value which is the resultant of the collector voltage of the transistors Tr 14  and Tr 15  in their conductive states and the forward voltage of the diodes D 1  and D 3 . Therefore, the voltage at the cathode (i.e., at point H in FIG. 6B) of the diode D 2  equals the resultant collector voltage of the transistors Tr 14  and Tr 15 . This collector voltage is fed to the base of the transistor Tr 13  to regulate the collector current of the latter. Then, the collector current of the transistor Tr 13  flows through the resistor R 1  to control the base voltage of the transistor Tr 10 . The collector current of the latter transistor is determined in accordance with its base voltage. Then, the collector current of the transistor Tr 10  flows as the base current I B  for the switching transistors Tr 14  and Tr 15  when the transistors Tr 11  and Tr 12  are alternately made conductive. 
     For example, if the respective base currents of the transistors Tr 14  and Tr 15  exceed the current necessary to drive the same to their proper saturation levels, the voltage at the point H becomes so low that the collector current of the transistor Tr 13  is reduced. As a result, the collector current of the transistor Tr 10  is also reduced. Consequently, the base currents I B  of the switching transistors Tr 14  and Tr 15  are reduced. By contrast, if the base current I B  is insufficient, the diodes D 1  -D 3  and the transistors Tr 13  and Tr 10  act to increase the base currents I B  for the transistors Tr 14  and Tr 15 . An equilibrium condition is quickly attained, in which the collector voltages of the transistors Tr 14  and Tr 15  at the point H substantially equals the base-emitter voltage (i.e., a level of about 0.7 V) of the transistor Tr 13 . In such an equilibrium condition, the collector-emitter voltages of the switching transistors Tr 14  and Tr 15  takes on a value of about 0.7 V, which is very close to the substantially saturated voltage. 
     In other words, even if the amplitude of the current to be fed to the coils 5A, 5A&#39;, 5B and 5B&#39; is changed in dependence upon the intensity of the supply voltage V s  fed to the coils 5A, 5A&#39;, 5B and 5B&#39;, only the minimum base current necessary for saturating the transistors Tr 14  and Tr 15  flows to their respective bases. Consequently, a high base current necessary for starting the motor can be provided, and when the motor attains its properly rated rotational speed, the base current is automatically reduced to the minimum necessary level. Thus, the power consumed by the motor can be further reduced. 
     In the present embodiment, moreover, the respective energizing periods of the A-phase stator coils 5A and 5A&#39; and the B-phase stator coils 5B and 5B&#39; are made to overlap each other slightly, just before and just after the current change-over time, as in the FIG. 3 embodiment. More specifically, the AC output signals, as shown in FIG. 7A from the output terminals A and B of the Hall element 10 are supplied to the bases of a pair of transistors Tr 16  and Tr 17  arranged as a differential amplifier, so that amplified output voltages, as shown by g and h in FIG. 7A, are generated from the collectors of these differentially-arranged transistors Tr 16  and Tr 17 . The resultant output voltages are supplied to the bases of another pair transistors Tr 18  and Tr 19 , respectively. Since these transistors Tr 18  and Tr 19  are biased to have an ON level (i.e., a threshold level), as shown in FIG. 7A, a pair of pulse signals i and j having high level sections overlapping each other, as shown in FIGS. 7C and 7D, respectively, are generated from the collectors of those transistors Tr 18  and Tr 19 . Such pulse signals signals i and j take on high levels when the signals h and g drop below their respective threshold levels. 
     A pair of diodes D 4  and D 5  have their respective cathodes connected to the collectors of the transistors Tr 19  and Tr 18 . The anodes of the diodes D 5  and D 6  are coupled to the base of a transistor Tr 20 . A biasing resistor is connected to the base of the transistor Tr 20 , while collector resistors R 6  and R 7  are respectively coupled to the collectors of the transistors Tr 18  and Tr 19 . 
     Since the cathode potentials of the diodes D 4  and D 5  are raised while both the pulse signals i and j take on high levels, no current flows at such times through a path consisting of the resistor R 5 , the diode D 5 , the resistor R 6 , the resistor R 7 , and the diode D 4 . As a result, the base potential of the transistor Tr 20  is increased in the neighborhood of the stator current change-over time, as shown in FIG. 7E, and this transistor Tr 20  is rendered non-conductive just before and just after such change-over times. 
     Another transistor Tr 21  has its base coupled to the collector of the transistor Tr 20  and is accordingly rendered conductive in the neighborhood of the stator current change-over time. Since the collector potential of the transistor Tr 21  rises when it is rendered conductive, a pair of overlap switching transistors Tr 22  and Tr 23 , whose bases are coupled to the collector of the transistor Tr 21 , are made conductive thereby. These transistors Tr 22  and Tr 23  are respectively coupled, through the collector resistors R 8  and R 9 , to the stator coils 5B, 5B&#39; and 5A, 5A&#39;. 
     If, for example, the A-phase stator coils 5A and 5A&#39; are energized by the current change-over signal f shown in FIG. 7B, a torque-compensating current for compensating the torque ripple flows through the B-phase stator coils 5B and 5B&#39; in the neighborhood of the current change-over time. Correspondingly, if the B-phase stator coils 5B and 5B&#39; are energized, the torque compensating current flows through the A-phase stator coils 5A and 5A&#39;. 
     With the drive circuit of this embodiment, the drop of the torque at the current change-over time can be compensated by feeding the compensating current through the overlap energizing transistor Tr 22  or Tr 23  to the stator coils 5A and 5A&#39; or 5B and 5B&#39; which would otherwise not be conducting. Thereby, the torque characteristics of the motor are substantially flattened. 
     Incidentally, the compensation for the torque drop can be set at any desired level by adjusting the collector resistors R 8  and R 9  of the transistors Tr 22  and Tr 23 . 
     The description thus far is directed to the circuitry of FIGS. 6A and 6B when operated in a forward rotational mode. At this time, the supply reel motor generates the torque opposite to the direction of translation of the magnetic tape and thus, owing to the low torque ripple of the motor, provides a remarkably even back tension to the magnetic tape. 
     The operation of the circuit for reverse rotation of the motor will now be described. 
     To achieve reverse operation the switch SW is switched from the rotational speed controlling voltage terminal V S  to the reversing voltage terminal V R . Then, a voltage at a low level (e.g., ground level) is fed from terminal 20 through the aforementioned terminal V R  to one terminal of each of the stator coils 5A and 5A&#39;, and 5B and 5B&#39;. It should be remarked that in the fast-forward mode the supply reel motor rotates in its reverse direction, corresponding to the direction of translation of the magnetic tape, whereby the tape running load is lightened. 
     The low voltage at the terminal 20 is also fed to the base of the transistor Tr 9  so that this transistor Tr 9  is rendered non-conductive. 
     For driving the motor in the reverse direction, the drive circuit includes a pair of switching transistors Tr 24  and Tr 25  whose collectors are coupled together to the power source V CC  and whose collectors are respectively coupled through diodes D 6  and D 7  to the stator coils 5A, 5A&#39; and 5B, 5B&#39;. The bases of the switching transistors Tr 24  and Tr 25  are respectively coupled to the collectors of the transistors Tr 7  and Tr 8 . 
     When, during a reverse operation, the transistor Tr 7  (or Tr 8 ) is turned on, the collector potential of the transistor Tr 7  (or Tr 8 ) is raised so that the potential at point I (or J) in FIG. 6A is raised to render the transistor Tr 24  (or Tr 25 ) conductive. In such a fashion, the transistor Tr 24  and Tr 25  are alternately turned on and off at equal intervals in dependence upon the aforementioned current change-over signal f. 
     It will be understood that in the reverse rotational mode, current passes from the power source V cc  through the A-phase stator coils 5A and 5A&#39; and the B-phase stator coils 5B and 5B&#39; in the direction opposite to that in the forward rotational mode. As a result, the motor is driven in the backward direction. 
     In this case, since the transistor Tr 9  is non-conductive, the transistors Tr 13  and Tr 10  are both non-conductive as are the transistors Tr 11  and Tr 12 . As a result, since no base current I B  is supplied to the transistors Tr 14  and Tr 15 , the current for forward drive, as has been described before, is not supplied to the A-phase and B-phase stator coils 5A, 5A&#39; and 5B, 5B&#39;. In the forward rotational mode, on the other hand, the aforementioned pair of transistors Tr 24  and Tr 25  are both held non-conductive so that no current for reverse drive will flow. 
     As further shown in FIG. 6A, a reference switching transistor Tr 26  is coupled to the base of the transistor Tr 20  and has its base coupled through diodes D 8  and D 9  to the stator coils 5A, 5A&#39; and 5B, 5B&#39;. Thus, when a reverse rotational mode is selected, the transistor Tr 26  is operative to render both the overlap energizing transistors Tr 22  and Tr 23  non-conductive. Consequently, when the supply reel motor is switched to its rewinding mode (or the take-up reel motor is switched to its fast forward mode) so that a voltage at a high level is applied to the terminal 20 and the motor is in its reverse rotational mode, the transistors Tr 22  and Tr 23  are prevented from heating. However, when the motor torque is used to apply back tension upon the supply reel in the recording or reproducing mode, the transistor Tr 26  is left non-conductive, and the overlap energizing transistors Tr 22  and Tr 23  are alternately turned on and off, as in the foregoing description, thereby providing compensation for torque drop. 
     On the contrary, in the high-speed forward rotation such as during a fast-forward operation, potentials at the points K and L take on high levels. The voltages at the points K and L are detected by means of the diodes D 8  and D 9  and the transistor Tr 26  is switched to its conductive state in response to such detected voltage. As a result, the transistor Tr 20  is turned on, whereas the transistor Tr 21  is turned off. Since, as a consequence, the overlap energizing transistors Tr 22  and Tr 23  are rendered inoperative, the heat liberated therein is reduced. While no torque-ripple correction is performed, torque ripple in the motor does not present any significant problem in the rewind and fast-forward operations. 
     FIGS. 8 and 9A-E show a motor driving circuit according to a third embodiment of this invention. Elements of this embodiment in common with those appearing in the first two embodiments are identified with the same reference characters, and a detailed explanation thereof will not be repeated. 
     In this embodiment, as is apparent from FIG. 8, there are not provided the separate overlap energizing transistors Tr 3  and Tr 4  and the torque compensation adjusting resistors R 8  and R 9  which are used in the first embodiment. Here, the AC output signals A 1  and B 1  (FIG. 9B) from the output terminals A and B of the Hall element 10 are fed to inputs of each of a pair of differential amplifiers 13 and 14. The differential amplifiers 13 and 14 each have a differential output coupled to the respective bases of switching transistors Tr 27  and Tr 28  arranged for energizing the stator coils 5A, 5A&#39; and 5B, 5B&#39;, respectively. 
     In this embodiment, the transistors Tr 27  and Tr 28  are each turned on for a period exceeding an electrical angle of 180°, in response to the output signals generated by the differential amplifiers 13 and 14. The A-phase and B-phase stator coils 5A and 5A&#39;, and 5B and 5B&#39; are alternately energized in an overlapping manner. 
     In this embodiment, the torque curves of the A-phase and B-phase stator coils intersect at the position of substantially half of peak torque as shown in FIG. 9A to take on a relationship of α=β, where α is the difference in torque from the torque at the points of intersection to maximum torque, while β is the amount of torque at the points of intersection. This relation α=β is achieved either by changing the magnetizing patterns of the magnets 2a, 2b, 3a and 3b of FIG. 1 or by changing the pitch angle (i.e., the angular spacing between the forward path portion V and the backward path portion U) of the stator coils 5A, 5A&#39;, 5B and 5B&#39;. Similarly to the first embodiment, pulse signals c and d, as shown in FIGS. 9C and 9D, are generated from the differential amplifiers 13 and 14 so that the transistors Tr 27  and Tr 28  are both rendered conductive over a period t&#39;, as shown in FIG. 9E. In other words, those transistors Tr 27  and Tr 28   supply current for an electrical angle of 180°+γ, where γ is a positive angle. As a result, an overlap energizing operation is performed over the aforementioned period t&#39; so that the torque drop at the current change-over time can be ideally compensated to the same level as that of the flat portion of the A-phase and B-phase torques, as indicated by chain lines in FIG. 9A. In this embodiment, this compensation is achieved by the simultaneous application of current to the A-phase and to the B-phase stator coils 5A, 5A&#39;, and 5B, 5B&#39;. 
     With the construction of this embodiment, because the resistors R 8  and R 9  are omitted, very little heat is liberated, and the driving circuit is highly suited to be formed as an integrated circuit. 
     Although several particular embodiments of this invention have been described in detail herein with reference to the accompanying drawings, it is to be understood that the invention is not limited to those precise embodiments and that various changes and modifications may be effected therein by persons skilled in the art without departing from the scope of spirit of the invention as defined in the appended claims.