Abstract:
A radio transmitter for multi-standard mobile communication systems has two stages of frequency up-conversion, the first being a digital process generating a variable IF and the second being an analogue conversion using a fixed frequency local oscillator, which alleviates the need for RF filtering after the final stage of power amplification. The IF frequency band is symmetrical about zero frequency and thus includes negative frequencies.

Description:
FIELD OF THE INVENTION 
     The present invention relates to a radio transmitter having particular, but not exclusive, application in digital communication systems such as GSM. 
     BACKGROUND OF THE INVENTION 
     The GSM specification for spurious emissions from a mobile station transmitter, as defined in the European Telecommunication Standards Institute (ETSI) document “GSM: Digital Cellular Telecommunications System (Phase 2) Radio Transmission and Receptions (GSM 05.05 version 4.19.0)”, is summarised in FIG. 1 of the accompanying drawings. The figure plots the permitted levels of unwanted noise N in a 1 Hz bandwidth, referenced to a carrier power level of +33 dBm at 902 MHz, against frequency f in MHz. Also shown are the positions of the GSM transmit (Tx) and receive (Rx) bands. In the portion of the GSM receive band between 925 and 935 MHz the noise must be held below −150 dBc, and in the portion between 935 and 960 MHz the noise must be held below −162 dBc. Such low levels of noise are difficult to achieve with a fully-integrated transmitter, and to meet this specification with a conventional architecture it is necessary to use an expensive RF filter after the final stage of power amplification, with a consequent loss of transmitter efficiency. 
     A block diagram of a conventional transmitter architecture, which performs dual up-conversion in analogue circuitry, is shown in FIG. 2 of the accompanying drawings. Digital data for transmission is provided as an input  202  to a Gaussian Minimum Shift Keying (GMSK) modulator  204  which produces as output analogue in-phase I and quadrature phase Q signals on a zero-frequency carrier. The I signal is supplied to a first IF mixer  206 , and the Q signal is supplied to a second IF mixer  208 . An output signal from a first Voltage Controlled Oscillator (VCO)  210  is supplied via a first 90° phase shifter  212  to the local oscillator port of the first IF mixer  206 , and directly to the local oscillator port of the second IF mixer  208 . The resultant output signals from the mixers  206 ,  208  are added together by a combiner  214  and filtered in a bandpass filter  222  to produce a signal at the required IF frequency, for example 100 MHz. As well as removing unwanted mixing products the bandpass IF filter  222  reduces levels of out-of-band noise. The filter  222  is commonly implemented off-chip. 
     The first VCO  210  is driven by a signal produced by an IF synthesiser  216  which derives its output using a 13 MHz reference oscillator  218  under the control of instructions passed on a control bus  220  to produce a fixed IF output. 
     The filtered IF signal is split into two parts. The first part has its phase shifted 90° by a second phase shifter  224  and is then up-converted by a first RF mixer  226 , the second part of the IF signal is up-converted by a second RF mixer  228 . An output signal from a second VCO  230  is supplied directly to the local oscillator port of the second RF mixer  228 , and via a third 90° phase shifter  232  to the local oscillator port of the first RF mixer  226 . The resultant output signals from the mixers  226 ,  228  are added together by a combiner  234  to produce a combined RF signal including a product at the required frequency in the GSM transmit band between 880 and 915 MHz. 
     The second VCO  230  is driven by a signal produced by an RF synthesiser  236  which derives its output using a 13 MHz reference oscillator  218  under the control of instructions passed on a control bus  220  to produce a variable output frequency. 
     Without extra filtering, noise from the second VCO  230  would fall into the GSM receive band at an unacceptably high level. The RF signal therefore passes through a first RF bandpass filter  238  before being amplified for transmission by a power amplifier  240 . The amplifier  240  is normally operated under heavy compression for best efficiency, and this has the effect of removing the AM component of single-sideband noise on the input signal. Without the AM component, the residual FM component produces noise at equal levels on the two sides of the signal, largely restoring noise in the unwanted sideband. Hence the signal must be filtered by a second RF bandpass filter  242  before transmission via an antenna  244 . The second RF filter  242  is much less desirable than the first filter  238  both in terms of cost (because of the higher power levels it must handle) and because of the resultant loss in transmitter power due to losses in the filter  242 . These transmitter losses can amount to more than 1 W, requiring the use of a bigger power amplifier  240  and a larger battery. 
     Such an architecture therefore has a number of disadvantages for use with current digital cellular communications standards. It is difficult to use for telephones operating in accordance with two or more standards unless the schemes are compatible (in the sense of having similar requirements for bandwidth and modulation schemes, for example). This is because only the baseband circuitry is digital, and the analogue IF and RF circuitry is inherently less flexible. Also, as mentioned above, it is difficult to meet the GSM requirements for spurious emissions without additional filtering after the power amplification stage. 
     SUMMARY OF THE INVENTION 
     An object of the present invention is to address the problems described above. 
     According to the present invention there is provided a radio transmitter comprising modulation means for producing quadrature modulated signals, first frequency-translation means for translating said signals to a variable intermediate frequency (IF) signal in digital form, digital to analogue conversion means for converting said variable IF signal to analogue form, second frequency-translation means for translating said analogue IF signal by a fixed frequency to a radio frequency (RF) signal, and power amplifying means for amplifying said RF signal for transmission. 
     The present invention is based upon the recognition, not present in the prior art, that digital up-conversion to a variable IF provides a more flexible transmitter architecture that does not require expensive RF filtering. 
     An advantage of the described transmitter architecture is that it is extremely versatile, giving the possibility of changing modulation methods, frequencies, sampling rates or bandwidths to accommodate a variety of communication standards. 
     Advantageously, error correction means are provided between the first and second frequency translation means, to correct for the imbalance between in-phase and quadrature signals in the second frequency translation means. 
     Provision of such means enables automatic calibration of the transmitter during manufacture to take account of the imbalance between signal paths in the second frequency-translation means, after which calibration no further attention is required. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     Embodiments of the present invention will now be described, by way of example, with reference to the accompanying drawings, wherein: 
     FIG. 1 is a graph of frequency f, in MHz, versus noise, N, in dBc/Hz, illustrating the GSM specification for spurious emissions, as described above; 
     FIG. 2 is a block diagram of a conventional transmitter architecture, as described above; and 
     FIG. 3 is a block diagram of a transmitter architecture designed in accordance with the present invention. 
    
    
     In the drawings the same reference numerals have been used to indicate corresponding features. 
     DETAILED DESCRIPTION OF THE INVENTION 
     An embodiment of a transmitter architecture designed in accordance with the present invention is shown in FIG.  3 . The basic concept is to make a first up-conversion a variable frequency conversion in the digital domain and a second a fixed frequency conversion in the analogue domain. 
     Digital data is provided as input  202  to a digital GMSK modulator  302 . The output of the modulator  302  is digitised I and Q signals on a zero-IF carrier at the GSM bit rate of 270833 bits per second. The signals are then processed by a digital up-conversion and filtering block  304 , which mixes the signals up to a variable IF of between −17.5 and +17.5 MHz. This is a purely digital process incorporating the necessary digital filtering. The up-rotation I and Q signals are derived from an IF synthesiser  306 , which receives control signals via a control bus  308 . 
     The tuning range reflects that required in the transmitter output for the GSM transmitter band. The use of a symmetrical range extending into negative frequencies keeps the IF as low as possible to minimise power consumption. As a consequence of the use of negative frequencies it is essential to process complex signals throughout the transmitter chain. The sampling rate after up-conversion must be at least 35 MHz to avoid aliasing. 
     Following this first up-conversion the signals pass through a digital error correction module  310 , discussed in more detail below. After suitable error corrections have been applied the I and Q digital signals are converted into analogue form by first and second digital to analogue converters  312 ,  322  and then filtered by first and second analogue lowpass filters  314 ,  324 , having a bandwidth of approximately 17.5 MHz. The filters  314 ,  324  can be low Q devices, capable of being implemented as active devices on-chip. 
     The filtered signals are then translated directly to the transmitter output frequency, the I signal being mixed by a first mixer  316  with an output signal from a fixed-frequency VCO  318  via a 90° phase shifter  320  and the Q signal being mixed by a second mixer  326  with the output signal from the VCO  318 . Advantages of using a fixed frequency VCO  318  include: 
     the VCO  318  does not require a multi-step synthesiser; 
     the design of the 90° phase shifter  320  is considerably simplified; and 
     the balance of the mixers is easier to ensure with a fixed-frequency local oscillator. 
     The two RF signals are combined by an adder  328 , after which the combined signals are filtered in an RF bandpass filter  330  to eliminate mixer spurious responses and reduce the levels of wideband noise. The signal is amplified by a power amplifier  332  and relayed to an antenna  244  for transmission. If the quality of the RF filter  330  is sufficiently high no additional filtering after the power amplifier  332  will be required. 
     It may be possible to eliminate the RF filter  330  if the fixed-frequency VCO  318  has a much higher Q than a tunable version and thereby generates lower levels of noise with less DC power consumption. However, the improvement in noise is in close-to-carrier noise performance rather than the level of noise at large frequency offsets, which is dominated by DC power consumption. Therefore to ensure that the transmitter noise is not degraded by the VCO with no RF filter  330  it would be necessary to make the VCO a relatively high power device. 
     Another issue relating to eliminating the RF filter  330  is that of the noise associated with the mixers  316 ,  326 . If the mixers have a noise figure of nominally 10 dB, and if the noise floor for the modulated signals driving the mixers  316 ,  326  is close to the theoretical limit of −174 dBm/Hz, the equivalent noise of the mixers at a 20 MHz offset cannot be below approximately −164 dBm/Hz. This in turn means that in order to achieve the desired −162 dBc/Hz of the GSM specification without the RF filter  330  the signal drive to the mixers  316 ,  326  would have to be about 0 dBm. Such a high level drive is not desirable if the spurious responses are to be kept under control, which is also necessary. 
     Hence, for most applications the disadvantages of eliminating the RF filter  330  will outweigh the advantage of lower component count. 
     The function of the digital error correction block  310  is to compensate for imperfections in the analogue front-end mixers  316 ,  326 . As in any practical integrated circuit, processing imperfections will lead to slight imbalances of the I and Q signal paths through the two analogue mixers  316 ,  326 . A fixed frequency local oscillator  318  should have been minimised these imbalances, but such are the constraints placed on the transmitter by the GSM specification that some error correction facility is likely to be required. The error correction block  310  applies suitable corrections in real time in the digital domain. An automatic calibration would be performed at the end of the manufacturing process of the product incorporating the transmitter, in which the transmitter output would be optimised for minimum spurious outputs. Once the correction terms had been computed they would be down-loaded into static memory in the error correction block  310 . After this process the error correction circuitry could correct for the imbalances in the mixers  316 ,  326 , assuming these imperfections remained constant with time. 
     The fixed frequency local oscillator  318  also helps to address another important requirement, that of maintaining the balance of the two mixers  316 ,  326  at the Local Oscillator (LO) ports. These must be held to tight tolerances to avoid the breakthrough of the LO signal into the transmitted signal. 
     If local oscillator pulling is a difficulty for channels selected in the middle of the GSM frequency range, it would be possible to change the IF frequency range from one of −17.5 MHz to +17.5 MHz to one of approximately zero to 35 MHz. This would move the local oscillator frequency out of the range of the transmitter and alleviate any pulling. 
     The architecture described above is extremely versatile and, with the inclusion of the RF filter  330 , noise should not be a problem even without a duplexer filter at the antenna  244 . All the signal processing elements between the input data stream  202  and the output of the error correction block  310  are implemented digitally, either in dedicated hardware or in software. Hence it is comparatively easy to change the modulation method, the frequencies, the sampling rates or the bandwidths to accommodate different communication standards from GSM. If the functions are implemented entirely in software, true multi-mode operation becomes a realistic proposition. The predominance of digital circuitry also offers a better prospect for using CMOS instead of the more expensive BiCMOS integrated circuit technologies, and no off-chip IF filtering is required. 
     An important advantage of the architecture is that is able to accommodate both constant and non-constant envelope modulation schemes as a consequence of avoiding the use of a phase-locked VCO. This may be particularly important for GSM if there is any move towards a non-constant envelope modulation scheme for higher bit rate traffic. 
     The architecture has good potential for dual-band operation in both the 800/900 MHz frequency bands and the 1800/1900 MHz bands without the need for heavy duplication of RF components. It may only be necessary to add a second, fixed frequency VCO, an RF filter and appropriate switches. 
     Although the present invention has been described with reference to the GSM cellular telecommunications system it will be apparent that it is equally applicable to other telecommunications systems, whether cellular or not. 
     From reading the present disclosure, other modifications will be apparent to persons skilled in the art. Such modifications may involve other features which are already known in radio transmitters, and which may be used instead of or in addition to features already described herein. 
     In the present specification and claims the word “a” or “an” preceding an element does not exclude the presence of a plurality of such elements. Further, the word “comprising” does not exclude the presence of other elements or steps than those listed.