Abstract:
This technology provides a method (application) of an algorithm to facilitate the design of wideband operations of antennas, and the design of sleeve cage monopole and sleeve helix, units. The technology is of interest/commercial potential throughout the audio communications community.  
     Omnidirectional capabilities and enhanced wideband capabilities are two desirable features for the design of many antenna applications. Designing omnidirectional antennas with wideband capabilities requires rapid resolution of complex relationship among antenna components to yield an optimal system. The invention comprises the use of a genetic algorithm with fitness values for design factors expressed in terms to yield optimum combinations of at least two types of antennas.  
     Cage antennas are optimized via a genetic algorithm (GA) for operation over a wide band with low voltage standing wave ratio (VSWR). Numerical results are compared to those of other dual band and broadband antennas from the literature. Measured results for one cage antenna are presented.  
     Genetic algorithms and an integral equation solver are employed to determine the position and lengths of parasitic wires around a cage antenna in order to minimize voltage standing wave ratio (VSWR) over a band. The cage is replaced by a normal mode quadrifilar helix for height reduction and the parasites are re-optimized. Measurements of the input characteristics of these optimized structures are presented along with data obtained from solving the electric field integral equation.  
     Genetic algorithms (Y. Rahmat-Samii and E. Michielssen,  Electromagnetic Optimizations by Genetic Algorithms,  New York: John Wiley and Sons, Inc., 1999) are used here in conjunction with an integral equation solution technique to determine the placement of the parasitic wires around a driven cage. The cage may be replaced by a quadrifilar helix operating in the normal mode in order to shorten the antenna. Measurements of these optimized structures are included for verification of the bandwidth improvements.

Description:
PRIORITY CLAIM  
       [0001]    This application claims the benefit of previously filed U.S. Provisional Application with the same titles and inventors as present, assigned U.S. Ser. No. 60/215,434, filed on Jun. 30, 2000, and which is incorporated herein by reference. 
     
    
     
       INTRODUCTION  
         [0002]    This technology provides a method (application) of an algorithm to facilitate the design of wideband operations of antennas, and the design of sleeve cage monopole and sleeve helix, units. The technology is of interest/commercial potential throughout the audio communications community.  
           [0003]    Omnidirectional capabilities and enhanced wideband capabilities are two desirable features for the design of many antenna applications. Designing omnidirectional antennas with wideband capabilities requires rapid resolution of complex relationship among antenna components to yield an optimal system. The invention comprises the use of a genetic algorithm with fitness values for design factors expressed in terms to yield optimum combinations of at least two types of antennas.  
           [0004]    Cage antennas are optimized via a genetic algorithm (GA) for operation over a wide band with low voltage standing wave ratio (VSWR). Numerical results are compared to those of other dual band and broadband antennas from the literature. Measured results for one cage antenna are presented.  
           [0005]    Genetic algorithms and an integral equation solver are employed to determine the position and lengths of parasitic wires around a cage antenna in order to minimize voltage standing wave ratio (VSWR) over a band. The cage is replaced by a normal mode quadrifilar helix for height reduction and the parasites are re-optimized. Measurements of the input characteristics of these optimized structures are presented along with data obtained from solving the electric field integral equation.  
           [0006]    Genetic algorithms (Y. Rahmat-Samii and E. Michielssen,  Electromagnetic Optimizations by Genetic Algorithms,  New York: John Wiley and Sons, Inc., 1999) are used here in conjunction with an integral equation solution technique to determine the placement of the parasitic wires around a driven cage. The cage may be replaced by a quadrifilar helix operating in the normal mode in order to shorten the antenna. Measurements of these optimized structures are included for verification of the bandwidth improvements.  
         BACKGROUND AND SUMMARY OF THE INVENTION  
         [0007]    Recent advances in modem mobile communication systems, especially those which employ spread-spectrum techniques such as frequency hopping, require antennas which have omnidirectional radiation characteristics, are of low profile, and can be operated over a very wide frequency range. The simple whip and the helical antenna operating in its normal mode appear to be attractive for this application because they naturally have omnidirectional characteristics and are mechanically simple. However, these structures are inherently narrow band and fall short of needs in this regard. Hence, additional investigations must be undertaken to develop methods to meet the wide bandwidth requirement of the communication systems.  
           [0008]    This invention comprises a method to design (produce) a product and the product(s) designed/produced as a result of the application of the method. The products are broad band, omnidimensional communications antennas, and the design procedure involves the coordinated, sequential application of two algorithms: a generally described “genetic algorithm that simulates population response to selection and a new algorithm that is a fast wire integral equation solver that generates optimal multiple antenna designs from ranges of data that limit the end product. Individual designs comprise a population of designs upon which specified selection by limiting the genetic algorithm ultimate identifies the optimum design(s) for specified conditions. Superior designs so identified can be regrouped and a new population of designs generated for further selection/refinement.  
           [0009]    The products are the antenna designs and specifications derived as a product of the application of the method briefly described above. The antennas all are characterized generally as broad band and omni directional, two features of critical importance in antenna design. In addition, although much of the theory has been developed on monopole antennas, both the method and designs include both monopole and dipole designs. In addition, the designs include sleeve-cage and sleeve-helix designs as hereinbelow further described.  
           [0010]    The cage monopole comprises four vertical, straight wires connected in parallel and driven from a common stalk at the ground plane. The parallel straight wires are joined by crosses made of brass (or other conductive) strips, the width of which is equal to the electrical equivalent of the wire radius. Compared to a single wire, this cage structure has a lower peak voltage standing wave ratio (VSWR) over the band. A structure with lower VSWR is amenable to improved bandwidth characteristics with the addition of parasitic elements.  
           [0011]    Adding parasitic elements of equal height and distance from the center of the cage monopole creates a sleeve cage monopole. The sleeve cage monopole has a greater bandwidth than its otherwise comparable antennas. Fitness values are determined by relative bandwidth, with greater fitness being associated with wider bandwidth defined by f 2 /f 1 , where f 2  and f 1  are respectively the largest and smallest frequencies between which VSWR is 3.5 or less. Speed of optimization is increased by interpolation of the impedance matrix.  
           [0012]    The heart of the process is the solution of the equation governing total axial current. The executable algorithm linked to the genetic algorithm by the fast wire integral equation solver provides a rapid method of solving this equation for varied values and inputs. The basic theory and equations are incorporated completely herein. See, S. D. Rogers and C. M. Butler, “An efficient curved-wire integral equation solution technique,” submitted to  IEEE Trans. Antennas Propagat.    
           [0013]    Reduced height without loss of bandwidth or omni-directional capabilities is a desired feature of antenna designs for a plurality of applications. These include installations in vehicles and confined interior spaces. The helix structure yields shorter antennas than the traditional whip structure with otherwise comparable features. Height is a function of the pitch angle of the helix, such that a pitch angle of 42 degrees reduces height by 30 percent. The addition of parasitic elements reduces VSWR in a helix configuration in a magnitude similar to the reduction noted for the cage monopole design.  
           [0014]    Many modern wireless communication systems require low-profile antennas. To meet this requirement, we consider the helical antenna operating in the normal mode. A normal mode helix and a straight wire antenna having approximately the same wire length exhibit similar input impedance and far field patterns. One drawback to the helix operating in the normal mode is that its bandwidth is too limited for many applications. To increase the bandwidth, we have considered several potential remedies, one of which is discussed in this paper. It is well known that adding additional parasitic straight wires on either side of a driven dipole antenna may increase the bandwidth of the dipole (J. L. Wong and H. E. King, AP-21, no. 5, 725-727, Sept. 1973). One must be especially careful, however, to choose parasitic elements with the proper length and spacing. We use this basic idea to increase the bandwidth of the helical monopole. A structure similar to the sleeve dipole but applied to the helix has been used to design dual frequency antennas (P. Eratuuli, et. al., Electronics Letters, v. 32, no. 12, 1051-1052, June 1996). The helix and its helical sleeve are both driven in the antenna of this reference. Several novel antenna structures are considered such as a driven helical antenna adjacent to parasitic helices and straight wires. Another candidate structure consists of a driven helix with helical parasites inside or outside of the driven element, which has the added benefit of conserving space. In any case, due to the large number of parameters in a helix, it is more difficult to design a broadband sleeve helical antenna than is the case for a sleeve dipole. It is not feasible to obtain optimum values of parameters by trial and error. Thus we employ a genetic algorithm routine (D. L. Carroll, A FORTRAN Genetic Algorithm Driver, http://www.staff.uiuc.edu/˜carroll/ga.html) and efficient integral equation solution techniques to optimize the antenna system for bandwidth. Having an efficient numerical solution technique is necessary for this problem since the geometry of the antenna is redefined for each structure evaluated by the genetic algorithm. Since these antennas have a high degree of curvature, their solutions generally require a large number of unknowns for representing the geometry. An efficient solution technique which gets around this problem is used (S. D. Rogers and C. M. Butler, APS Symposium Digest, vol. I, 68-71, July 1997).  
           [0015]    1) Commission B. B-2 Antennas  
           [0016]    2) A genetic algorithm is used to optimize helical parasitic elements for a helical antenna.  
           [0017]    3) This work is an extension to increasing the bandwidth of dipoles by use of parasites. This research could not have been completed in a time efficient manner without the development of an efficient integral equation solution technique for curved wires with reference below.  
           [0018]    S. D. Rogers and C. M. Butler, “Reduced Rank Matrices for Curved Wire Structures,” Digest of IEEE APS Symposium, Montreal, Canada, vol. 1, pp. 68-71, July 1997.  
           [0019]    We have recently shown from numerical calculations that the bandwidth of a normal mode helix can be increased by the addition of close-by wire parasites (S. D. Rogers, J. C. Young, and C. M. Butler, “Bandwidth Enhanced Normal Mode Helical Antennas,”  Digest  1998  USNC/URSI National Radio Science Meeting,  Atlanta, Ga., p. 293, June 1998). A genetic algorithm and a fast integral equation solution technique are employed to determine the optimum distance and height of these parasites. In (H. E. King and J. L. Wong, “An Experimental Study of a Balun-Fed Open-Sleeve Dipole in Front of a Metallic Reflector,”  IEEE Trans. Antennas Propagat.  (Commun.), vol. AP-20, pp. 201-204, March 1972) these parameters were determined experimentally when the driven element was a straight wire dipole. In a recent paper (H. Nakano, et. al., “Realization of Dual-Frequency and Wide-Band VSWR Performances Using Normal-Mode Helical and Inverted-F Antennas,”  IEEE Trans. Antennas Propagat.,  vol. AP-46, June 1998) a central parasitic straight cylinder was added inside a driven single wire helix to obtain dual frequency operation. We have found that even greater bandwidth, over that of a single driven wire, can be realized when the parasites are placed around “cage” monopoles having several parallel wires. Similar observations are made about a multifilament versus a single filament helix.  
           [0020]    Bandwidth of vertically polarized wire antennas is often increased by adjustment of the antenna geometry. King and Wong reduce VSWR by placing parasitic wires around a driven element, creating the well-known open sleeve dipole (KING, H. E., and WONG, J. L.: ‘An experimental study of a balun-fed open-sleeve dipole in front of a metallic reflector’,  IEEE Trans. Antennas Propagat.,  March 1972, 20, (2), pp. 201-204). In (NAKANO, H., IKEDA, N., WU, Y., SUZUKI, R., MIMAKI, H., and YAMAUCHI, J.: ‘Realization of dual-frequency and wide-band VSWR performances using normal-mode helical and inverted-F antennas’,  IEEE Trans. Antennas Propagat.,  June 1998, 46, (6), pp. 788-793) the displacement of a parasitic monopole is varied inside a driven normal-mode helical antenna in order to control its characteristics. Cage antennas can be made broadband when their dimensions are chosen judiciously. Genetic algorithms and integral equation solution techniques are employed here to optimize the dimensions of the cage antenna in order to create a structure with low VSWR over a wide band.  
           [0021]    Recent advances in modem mobile communication systems, especially those which employ spread-spectrum techniques such as frequency hopping, require low-profile, broadband, omnidirectional (in azimuth) antennas. The simple whip and the helical antenna, operating in its normal mode, are potentially attractive for these applications because they naturally have suitable radiation characteristics and are mechanically simple and rugged. However, these structures are inherently narrow band. Hence, additional measures are employed to meet the wide bandwidth requirement of communication systems. Antennas often are loaded with tuning circuits and are connected to radios through matching networks in order to improve overall bandwidth. Altering the antenna geometry is another method for modifying bandwidth properties. The sleeve monopole, in which the outer conductor of the coaxial feed line forms a “sleeve” around the base of the protruding center conductor, is known to have greater bandwidth than the conventional monopole and has been studied extensively (J. Taylor, “The sleeve antenna,” doctoral dissertation, Cruft Lab., Harvard Univ., Cambridge, Mass., 1950); (R. W. P. King, The Theory of Linear Antennas. Cambridge, Mass.: Harvard Univ Press, 1956); (A. J. Poggio and P. E. Mayes, “Pattern bandwidth optimization of the sleeve monopole antenna,”  IEEE Trans. Antennas Propagat.  (Commun.), vol. AP-14, pp. 643-645, Sept. 1966); (Z. Shen and R. MacPhie, “Rigorous evaluation of the input impedance of a sleeve monopole by modal-expansion method,”  IEEE Trans. Antennas Propagat.,  vol. AP-44, pp. 1584-1591, Dec. 1996). A variation of this antenna is the open-sleeve dipole which has straight-wire parasites in place of the coaxial sleeve. The effects of the spacing and size of the parasitic elements on the VSWR are determined experimentally in (H. E. King and J. L. Wong, “An experimental study of a balun-fed open-sleeve dipole in front of a metallic reflector,”  IEEE Trans. Antennas Propagat.  (Commun.), vol. AP-20, pp. 201-204, March 1972). In other papers, parasitic and driven elements of various sorts are combined in order to create dual band antennas. In (P. Eratuuli, et. al., “Dual frequency wire antennas,”  Electronics Letters,  vol. 32, no. 12, pp. 1051-1052, June 6, 1996) the driven wire is a straight monopole or a helix surrounded by a parasitic helix. In (H. Nakano, et. al., “Realization of dual-frequency and wide-band VSWR performances using normal-mode helical and inverted-F antennas,”  IEEE Trans. Antennas Propagat.,  vol. AP-46, pp. 788-793, June 1998) the position of a straight-wire parasite inside a driven normal mode helical antenna is adjusted to control the VSWR over the band of operation. Another antenna, which can be made to have broadband properties if its dimensions are chosen judiciously, is the cage antenna (S. D. Rogers and C. M. Butler, “Cage antennas optimized for bandwidth,” submitted to  Electronics Letters,  April 2000). The cage is more amenable than a single straight wire to improvement in bandwidth when parasitic wires of appropriate size and spacing are added (S. D. Rogers and C. M. Butler, “The sleeve-cage monopole and sleeve helix for wideband operation,”  Digest of APS Symposium,  Orlando Fla., vol. 2, pp. 1308-1311, July 1999).  
           [0022]    We have found that the cage structure and multifilar helices are more amenable than single wire antennas to improvements in VSWR when parasitic wires are added. The helical configuration can be used to reduce the height of the antenna, but at the sacrifice of bandwidth. While the addition of the parasitic wires improves the overall bandwidth, the VSWR increases outside the design band. Fast integral equation solution techniques and optimization methods have been developed in the course of this work and have led to effective tools for designing broadband antennas.  
           [0023]    Certain exemplary attributes of the invention may relate to a method to create optimum design specifications for omni-directional, wide band antennas comprising the steps of:  
           [0024]    (a) loading software including a genetic algorithm and an executable algorithm that is a fast wire equation solver into a computer;  
           [0025]    (b) loading instructions into said computer specifying basic antenna design to be optimized;  
           [0026]    (c) loading antenna design parameters and corresponding ranges of values for said parameters into said computer;  
           [0027]    (d) specifying resolution of said parameters by loading number of bits per parameter into said computer;  
           [0028]    (e) executing (operating) said genetic algorithm thereby generating a population of individual antenna designs each with a fitness value; and  
           [0029]    (f) evaluating relative fitness of antenna designs produced and selecting superior designs for continued refinement.  
           [0030]    The foregoing method may further comprise the following exemplary subroutines and algorithms for the software involved:  
           [0031]    (a) a first algorithm that allows different values for critical design elements to combined in all possible combinations and a fitness value for each design ultimately estimated;  
           [0032]    (b) a second algorithm that determines electronic current in an antenna by solving an integral equation numerically;  
           [0033]    (c) a computer program link that provides essential communication between said first algorithm and said second algorithm.  
           [0034]    Certain exemplary attributes of the invention may further relate to the sleeve monopole antenna designs, the cage sleeve monopole antenna designs, and the sleeve dipole antenna designs produced following the foregoing methods. Those of ordinary skill in the art will appreciate that various modifications and variations may be practiced in particular embodiments of the subject invention in keeping with the broader principles of the invention disclosed herein. The disclosures of all the citations herein referenced are fully incorporated by reference to this disclosure.  
       
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0035]    Design Procedure  
         [0036]    We modeled and measured the properties of a so-called cage monopole. The cage monopole shown in FIG. 101 a.  consists of four vertical straight wires connected in parallel and driven from a common stalk at the ground plane. The ground plane in this model is assumed to be of infinite extent to facilitate analysis. The parallel straight wires are joined by crosses constructed of brass strips. The strip width was selected to be electrically equivalent to the wire radius for the purpose of modeling the structure. Compared to a single, thin, straight wire, the cage structure with multiple wires has a lower peak voltage standing wave ratio (VSWR) over the band. This is important since a structure which has a comparatively small VSWR over a band is more amenable to improvements in bandwidth with the addition of other components such as loads or parasites than is the common single-wire monopole with higher VSWR.  
         [0037]    Next we add four parasitic straight wires of equal height (h) and distance (r) from the center of the cage to create the so-called “sleeve-cage monopole” of FIG. 102 a.  The genetic algorithm of (D. L. Carroll, “A FORTRAN Genetic Algorithm Driver”, Univ. of Illinois, Urbana, Ill., http://www.staff.uiuc.edu/˜carroll/ga.html) is used to determine the optimum distance and height of these parasitic straight wires. In this example the fitness value assigned to each antenna in the optimization process is the bandwidth ratio defined by f 2 /f 1 , where f 2  and f 1  are, respectively, the largest and smallest frequencies between which the VSWR is 3.5 or less. We interpolate the impedance matrix with respect to frequency in order to increase the speed of the optimization process.  
         [0038]    To design antennas that are smaller, we turn our attention to the normal mode helix, since, for operation about a given frequency, it can be made shorter than the vertical whip by adjustment of the pitch angle. Also, we observe a decrease in the peak VSWR when additional filaments are added to the helix driven from a central straight wire. Generally, a normal mode helix will exhibit electrical properties similar to those of a straight wire having the same wire length, though the peak VSWR for the helix is usually greater. The quadrifilar helix of FIG. 103 a  whose height is 9.8 cm can be used in the same bands as a cage monopole of height about 14 cm. Thus, the total height of the antenna can be reduced by 30% with the 42° pitch angle. Parasitic straight wires of optimum height and distance are added to create what we call the “sleeve helical monopole” shown in FIG. 104 a.    
         [0039]    Results and Discussion  
         [0040]    As one can see from the VSWR data, good agreement is achieved between predictions computed by means of our numerical techniques and results measured on a model mounted over a large ground plane. The frequency range over which data are presented is dictated by the frequencies over which our ground plane is electrically large. The slight discrepancies in the computed and measured results are attributed to imprecision in the construction of the antennas. The predicted results of bandwidth and VSWR of each antenna are summarized in the table below.  
                                                                                         Frequency   Height   Width       Structure   VSWR   BW Ratio   Range (MHZ)   (cm)   (cm)                                Cage monopole   &lt;5.0   11.7:1   300-3500   17.2   2.2           &lt;3.5     3:1   950-2850       Sleeve-cage   &lt;5.0    5.2:1   315-1650   17.2   5       monopole   &lt;3.5    4.4:1   350-1550       Quadrifilar   &lt;5.0    5.8:1   475-2750   9.8   2       helix   &lt;3.5    1.6:1   500-800        Sleeve helix   &lt;5.0    3.9:1   475-850    9.8   6           &lt;3.5    3.5:1   500-1750                  
 
         [0041]    We point out that when the parasitic elements are added to each structure, the bandwidth ratio increases for the VSWR&lt;3.5 requirement. However, outside of this frequency range the VSWR is worse than that of the antenna without parasites. In other words, VSWR has, indeed, been improved markedly over the design range but at a sacrifice in performance outside the range, where presumably the antenna would not be operated. Also, notice that the deep nulls in the directivity at the horizon for the cage and the quadrifilar helix structures have been eliminated with the addition of the parasites. Thus the directivity is improved in the band where on the basis of VSWR this antenna is deemed operable, although there was no constraint on directivity specified in the objective finction.  
         [0042]    Cage Antennas Opimized for Bandwidth  
         [0043]    Design Method: The cage antenna is depicted in FIG. 201 where one sees four vertical wires joined to the feed and stabilized by thin brass strips of width w. The strips are treated as wires of radius a=w/4. The GA of (CARROLL, D. L.: ‘Chemical Laser Modeling with Genetic Algorithms’,  AIAA Journal,  Feb. 1996, 34, (2), pp. 338-346) is applied to optimize the diameter (d) of the cage structure and the length (h 2 ) of the wires in the cage. Each function evaluation consists of numerically solving the electric field integral equation for the cage geometry (having dimensions chosen by the GA) over the band of interest. Candidate antennas are given a fitness score equal to the bandwidth ratio f h /f l  where f h  is the lowest and f h  is the highest frequency of operation over a band where the VSWR meets the design goal.  
         [0044]    Results: The antenna of FIG. 202 is optimized for a design goal of VSWR&lt;2.0 over the frequency band 500 to 1600 MHz. The GA picks the parameter d from a range of 1 cm to 5 cm with a resolution of 0.13 cm (5 bits, 32 possibilities). The range specified for parameter h 2  is 8 cm to 12 cm with a resolution of 0.27 cm (4 bits, 16 possibilities). The GA converges to an optimum solution after three generations with five antennas per generation. A sensitivity analysis reveals that antenna input characteristics change only modestly with small geometric variation. The directivity of this cage antenna for φ=0° and θ=75°, 90° is above 4 dBi over the entire band. The properties of this antenna and those of Nakano&#39;s helical monopole (NAKANO, H., IKEDA, N., WU, Y., SUZUKI, R., MIMAKI, H., and YAMAUCHI, J.: ‘Realization of dual-frequency and wide-band VSWR performances using normal-mode helical and inverted-F antennas’,  IEEE Trans. Antennas Propagat.,  June 1998, 46, (6), pp. 788-793), which is designed to operate with VSWR &lt;2.0 in two frequency bands, are listed in Table 201 for comparison.  
         [0045]    The antenna of FIG. 203 is optimized for a design goal of VSWR &lt;2.5 in the frequency range 200 to 1200 MHz. This range is chosen for comparison of the cage antenna to the open sleeve dipole of (KING, H. E., and WONG, J. L.: ‘An experimental study of a balun-fed open-sleeve dipole in front of a metallic reflector’,  IEEE Trans. Antennas Propagat.,  March 1972, 20, (2), pp. 201-204) which operates over the frequency range 225 to 400 MHz. The GA is allowed to chose parameter d from 1 cm to 10 cm with a resolution of 0.6 cm (4 bits, 16 possibilities). The parameter h 2  is selected from 20 cm to 25 cm with a resolution of 0.33 cm (4 bits, 16 possibilities). An optimum result is reached after 11 generations with five antennas per generation. This cage monopole is not useful over the entire frequency range for which its VSWR is less than 2.5 since there is a null in the directivity within this range. It is operable over a 3.6:1 bandwidth for VSWR less than 2.5 and directivity greater than 0 dBi. In Table 201 are listed the properties of the cage antenna together with those of the sleeve dipole.  
         [0046]    Cage Monopole an Sleeve-cage Monopole  
         [0047]    The cage monopole shown in FIG. 301 a  consists of four vertical straight wires connected in parallel and driven from a common wire which is the extension of the center conductor of a coaxial cable protruding from the ground plane. The ground plane in this model is assumed to be of infinite extent in the analysis of the structure. The parallel straight wires are joined by crosses constructed of brass strips. The strip width w was selected to be electrically equivalent to the wire radius a for the purpose of modeling the structure (w=4a ) (C. M. Butler, “The equivalent radius of a narrow conducting strip,”  IEEE Trans. Antennas Propagat.,  vol. AP-30, pp.755-758, July 1982). Compared to a single, thin, straight wire, the cage structure with multiple wires has a lower peak VSWR over the band as seen in FIG. 301 b.  This is important since a structure which has a comparatively small VSWR over a band is more amenable to improvements in bandwidth with the addition of other components such as loads or parasites than is the common single-wire monopole with higher VSWR.  
         [0048]    Four parasitic straight wires of equal height (h) and radial distance (r) from the center line of the cage are added to create the so-called “sleeve-cage monopole” of FIG. 302 a.  The genetic algorithm of (D. L. Carroll, “Chemical Laser Modeling with Genetic Algorithms,”  AIAA Journal,  vol. 34, no. 2), pp. 338-346, Feb. 1996) is used to determine optimum values of h and r for given design goals. An objective function evaluation for one antenna in the GA population involves numerically solving the electric field integral equation for many frequencies within the band of interest. Since this must be done for many candidate antennas, it is advantageous to interpolate the integral equation impedance matrix elements with respect to frequency (E. H. Newman, “Generation of wide-band data from the method of moments by interpolating the impedance matrix,”  IEEE Trans. Antennas Propagat.,  vol. AP-36, pp. 1820-1824, Dec. 1988). Each candidate structure is assigned a fitness value based on its electrical properties. A simple fitness value used here is the antenna bandwidth ratio which measures the performance of the antenna over a frequency band of interest denoted by [f A , f B ]. The bandwidth ratio for a particular antenna is considered a function of its geometry and is computed from  
         F        (     h   ,   r     )       =       f   2       f   1               where             f   1     =           min        (   f   )         f   ∈     [       f   A     ,     f   B       ]                         such  that                     VSWR        (   f   )         ≤     limit                   and                     f   2       =             max        (   f   )         f   ∈     [       f   1     ,     f   B       ]                         such  that                     VSWR        (   f   )         ≤       limit  for  all                   f       ∈       [       f   1     ,     f   2       ]     .                             
 
         [0049]    Another viable fitness value is the percent bandwidth defined here as  
         %      BW     =     100                         f   2     -     f   1             f   2          f   1           .                             
 
         [0050]    Quadrifilar Helix and Sleeve Helix  
         [0051]    To design low profile antennas, we turn our attention to the normal mode helix, since, for operation about a given frequency, it can be made shorter than the vertical whip by adjustment of the helix pitch angle. Generally, a normal mode helix will exhibit electrical properties similar to those of a straight wire having the same wire length, though the peak VSWR for the helix is usually greater. The helix exhibits vertical polarization as long as it operates in the normal mode. There is a decrease in the peak VSWR, relative to that of a single-wire helix, when additional helical filaments are added to one driven from a central straight wire. The quadrifilar helix of FIG. 303 whose height is 9.8 cm can be used in the same bands as a cage monopole of height about 14 cm. Thus, the total height of the antenna can be reduced by 30% with the 42° pitch angle. Parasitic straight wires of optimum height and distance are added to create what we call the “sleeve helical monopole” shown in FIG. 304. Most integral equation solution techniques for the helix are, in general, more computationally expensive since these require many basis functions to represent the vector direction of the current along the meandering wire. A solution procedure which uncouples the representation of the geometry from the representation of the unknown current is developed in (S. D. Rogers and C. M. Butler, “An efficient curved-wire integral equation solution technique,” submitted to  IEEE Trans. Antennas Propagat .) and is used here to reduce the time in optimization of antennas with curved wires.  
         [0052]    Results  
         [0053]    As one can see from the VSWR data, good agreement is achieved between predictions computed by means of numerical techniques (S. D. Rogers and C. M. Butler, “An efficient curved-wire integral equation solution technique,” submitted to  IEEE Trans. Antennas Propagat ) and results measured on a model mounted over a large ground plane. The frequency range over which our experiments are conducted is dictated by the frequencies over which the ground plane is electrically large. Of course, the dimensions of the antenna may be scaled for use in other bands. The slight discrepancies in the computed and measured results are attributed to the difficulty in building the antenna to precise dimensions. However, a sensitivity analysis reveals that the antenna performance changes minimally with small variations in geometry. The reflection coefficient is measured at the input of the coaxial cable driving the monopoles and of a shorted section of coaxial line having the same length. Applying basic transmission line theory to these data, one can determine the measured input impedance of the antenna with the reference “at the ground plane.” All VSWR data is for a 50Ω system. As the feed point properties of the various antennas are evaluated, we must also keep in mind the radiation properties of the antenna, so computed directivity is included herein. The predicted results of bandwidth and VSWR of each antenna are summarized in Table 301.  
                                                                           TABLE 301                           Summary of results.                            Frequency                               Range   Height   Width       Structure   VSWR   BW Ratio   BW %   (MHz)   (cm)   (cm)                    Cage   &lt;5.0   11.7   312   300-3500   17.2   2.2       monopole   &lt;3.5   3   115   950-2850       Sleeve-   &lt;5.0   5.2   185   315-1650   17.2   5       cage       monopole   &lt;3.5   4.4   163   350-1550       Quadri   &lt;5.0   5.8   199   475-2750   9.8   2       filar       helix   &lt;3.5   1.6   47   500-800       Sleeve   &lt;5.0   3.9   147   475-1850   9.8   6       helix   &lt;3.5   3.5   134   500-1750                  
 
         [0054]    We point out that, when the parasitic elements are added to each structure, the bandwidth ratio increases for the VSWR&lt;3.5 requirement. However, outside of this frequency range the VSWR is worse than that of the antenna without parasites. In other words, VSWR has, indeed, been improved markedly over the design range but at a sacrifice in performance outside the range, where presumably the antenna would not be operated. Also, notice that the deep nulls in the directivity at the horizon for the cage and the quadrifilar helix structures have been eliminated with the addition of the parasites. Thus the directivity is improved in the band where, on the basis of VSWR, this antenna is deemed operable, although there was no constraint on directivity specified in the objective function.  
         [0055]    The following is a detailed description (including documentary references) of an exemplary efficient curved-wire integral equation solution technique as may be practiced in accordance with the subject invention.  
         [0056]    ABSTRACT  
         [0057]    Computation of currents on curved wires by integral equation methods is often inefficient when the structure is tortuous but the length of wire is not large relative to wavelength at the frequency of operation. The number of terms needed in an accurate piecewise straight model of a highly curved wire can be large yet, if the total length of wire is small relative to wavelength, the current can be accurately represented by a simple linear function. In this paper a new solution method for the curved-wire integral equation is introduced. It is amenable to uncoupling of the number of segments required to accurately model the wire structure from the number of basis functions needed to represent the current. This feature lends itself to high efficiency. The principles set forth can be used to improve the efficiency of most solution techniques applied to the curved-wire integral equation. New composite basis and testing functions are defined and constructed as linear combinations of other commonly used basis and testing functions. We show how the composite basis and testing functions can lead to a reduced-rank matrix which can be computed via a transformation of a system matrix created from traditional basis and testing functions. Supporting data demonstrate the accuracy of the technique and its effectiveness in decreasing matrix rank and solution time for curved-wire structures.  
         [0058]    I. Introduction  
         [0059]    Numerical techniques for solving curved-wire integral equations [1] may involve large matrices, often due primarily to the resources needed to model the structure geometry rather than due to the number of basis functions needed to represent the unknown current. This is obviously true when a subdomain model is used to approximate a curvilinear structure in which the total wire length is small compared to the wavelength at the frequency of operation. Usually the number of segments needed in such a model is dictated by the structure curvature rather than by the number of weighted basis functions needed in the solution method to represent the unknown current. There is a demand for a general solution technique in which the number of unknowns needed to accurately represent the current is unrelated to the number of straight segments required to model (approximately) the meandering contour of the wire and the vector direction of the current. In recent years attention in the literature has been given to improving the numerical efficiency of integral equation methods for curved-wire structures [2]-[13]. For the most part, presently available techniques incorporate basis functions defined on circular or curved wire segments. The authors of [2] define basis and testing functions along piecewise quadratic wire segments and achieve good results with fewer unknowns than would be needed in a piecewise straight model of a wire loop and of an Archimedian spiral antenna. Others introduce solution techniques for structures comprising circular segments that numerically model the current specifically on circular loop antennas [3], [4]. An analysis of general wire loops is presented in [5], where a Galerkin technique is employed over a parametric representation of a superquadric curve. In [6] arcs of constant radii are employed to define the geometry of arbitrarily shaped antennas from which is developed a technique for analyzing helical antennas. Other methods which utilize curved segments for subdomain basis and testing functions are available [7]-[10].  
         [0060]    There are several advantages inherent in techniques in which basis and testing functions are defined over curved wire segments. Geometry modeling error can be made small and solution efficiency can be increased since to “fit” some structural geometries fewer curved segments are needed than is feasible with straight segments. Although these techniques are successful, they suffer disadvantages as well. First, the integral equation solution technique must be formulated to account for the new curved-segment basis and testing functions. This means that computer codes must be written to take advantage of the numerical efficiency of these new formulations incorporating the curved elements. A second disadvantage of curvilinear basis function modeling is that they fit one class of curve very well but are not well suited to structures comprising wires of mixed curvature. That is, circles fit loops and helices well but not spirals. Clearly, when a given structure comprises several arcs of different curvatures, the efficiency of methods employing a single curved-segment representation suffers. Elements like the quadratic segment or the arc-of-constant-radius segment increase the complexity of modeling. The third disadvantage of these techniques is that, for many structures, they do not lead to complete uncoupling of the number of the unknown current coefficients from the number of segments needed to model the structure geometry. For example, several quadratic segments or arcs, with one weighted unknown defined on each, would be required to model the geometry of one turn of a multitum helix, yet the current itself may be represented accurately in many cases by a simple linear function over several turns.  
         [0061]    In this paper, an efficient method for solving for currents induced on curved-wire structures is presented. The solution method is based on modeling the curved wire by piecewise-straight segments but the underlying principles are general and can be exploited in conjunction with solution procedures which depend upon other geometry representations, including those that use arcs or curves. It is ideal for multi-curvature wire structures [12], [13]. The improved solution technique depends upon new basis and testing functions which are defined over more than two contiguous straight-wire segments. Composite basis functions are created as sums of weighted piecewise linear functions on wire segments, and composite testing functions compatible with the new basis functions are developed. The new technique allows one to reduce the rank of the traditional impedance matrix. We show how the matrix elements for a reduced-rank matrix can be computed from the matrix elements associated with a traditional integral equation solution method. Of paramount importance is the fact that the number of elements employed to model the geometric features of the structure is unrelated to the number of unknowns needed to accurately represent the wire current.  
         [0062]    The concept of creating a new basis function as a linear combination of other basis functions is used in [14] for a multilevel iterative solution procedure for integral equations. Perhaps the composite basis function defined herein can be thought of as a “coarse level” basis function in multilevel terminology, although the method described in this paper is not related to the so-called multilevel or multigrid theory of [14]-[16].  
         [0063]    The improved solution technique requires fewer unknowns than the traditional solution to represent the current on an Archimedian spiral antenna. Results comparable to those presented in [2] are achieved for the spiral. The improved technique also allows one to significantly reduce the number of unknowns required to solve for the current on wire helices. Specifically, the results of a convergence test show that the current on a helix can be modeled accurately with the same number of unknowns needed for a “similar” straight wire even though the helix has a large number of turns.  
         [0064]    II. Integral Equation for General Curved Wires  
         [0065]    In this section we present the integro-differential equation governing the electric current on a general three dimensional curved or bent wire. Examples are the wire loop, the helix, and the meander line shown in FIG. 1. The wire is assumed to be a perfect electrical conductor and to be thin which means that the radius is much smaller than the wavelength and the length of wire. Under these thin-wire conditions the current is taken to be axially directed, circumferentially invariant, and zero at free ends. The equation governing the total axial current I(s)ŝ on the thin curved wire is  
                   -   j                     η     4      π                 k            {         k   2            ∫   C            I        (     s   ′     )                s   ^     ′     ·     s   ^            K        (     s   ,     s   ′       )                 s   ′             +               s              ∫   C                      s   ′              I        (     s   ′     )            K        (     s   ,     s   ′       )                            s   ′               }       =       -       E   i          (   s   )         ·     s   ^         ,     s   ∈   C             (   1   )                               
 
         [0066]    in which C is the wire axis contour, s denotes the arc displacement along C from a reference to a point on the wire axis, and s is the unit vector tangent to C at this point. The positive sense of this vector is in the direction of increasing s. K(s,s′) is the kernel or Green&#39;s function,  
                 K        (     s   ,     s   ′       )       =       1     2      π              ∫     -   π     π                     -   j                   kR       R               φ   ′               ,           (   2   )                               
 
         [0067]    in which R is the distance between the source and observation points on the wire surface, and E i (s) is the incident electric field which illuminates the wire, evaluated in (1) on the wire surface at arc displacement s. Geometric parameters for an arbitrary curved wire are depicted in FIG. 2.  
         [0068]    III. Traditional Solution Technique  
         [0069]    The new solution method proposed in this paper can be viewed as an improvement to present methods. In fact, employing the ideas set forth in Section IV, one can modify an existing subdomain solution method to render it more efficient for solving the curved-wire integral equation. Hence, the new method is explained in this paper as an enhancement of a method that has proved useful for a number of years. The method selected for this purpose is based on modeling the curved wire as an ensemble of straight-wire segments, with the unknown current represented as a linear combination of triangle basis functions and testing done with pulses. In this section this method is outlined as a basis for the explanation of the new method in Section IV.  
         [0070]    The first step in modeling a curved wire is to select points on the wire axis and define vectors r 0 , r 1 , . . . , r p  from a reference origin to the selected points. The curved wire is modeled approximately as an ensemble of contiguous straight-wire segments joining these points (cf FIG. 3). The arc displacement along the axis of the piecewise linear approximation of C is measured from the reference point labeled r 0 . The arc displacement between r 0  and the n th  point located by r n , is l n . A general point on the piecewise-straight approximation of the wire axis is located alternatively by means of the vector r and by the arc displacement l from the reference to the point. Various geometrical parameters describing the wire can be expressed in terms of the vectors locating the points on the wire axis. The unit vectors along the directions of the segments adjacent to the point r p  shown in FIG. 4 are given by  
                 l   ^       p   -       =         r   p     -     r     p   -   1           Δ     p   -                 (   3   )                   l   ^       p   +       =         r     p   +   1       -     r   p         Δ     p   +                 (   4   )                               
 
         [0071]    The midpoint of the straight-wire segment joining r p  and r p±1  is located by  
               r     p   ±       =         1   2          [       r   p     +     r     p   ±   1         ]       .             (   7   )                               
 
         [0072]    In order to emphasize the fact that the model is now a straight wire segmentation of the original curved wire, s in (1) is replaced by l, the arc displacement along the axis of the straight wire model. With this notation and subject to the piecewise straight wire approximation, Eq. (1) becomes  
                   -   j                     η     4      π                 k            {         k   2            ∫   L            I        (     l   ′     )                l   ^     ′     ·     l   ^                       K        (     l   ,     l   ′       )                 l   ′             +               l              ∫   L                      l   ′              I        (     l   ′     )            K        (     l   ,     l   ′       )                 l   ′               }       =       -       E   i          (   l   )         ·     l   ^         ,     l   ∈   L             (   8   )                               
 
         [0073]    where L is the piecewise straight approximation to C.  
         [0074]    In a numerical solution of the integral equation for a curved wire structure, the (vector) current is expanded in a linear combination of weighted basis functions defined along the straight-wire segments. Even though they can be any of a number of functions, those employed here, for the purpose of illustration in this paper, are chosen to be triangle functions with support over two adjacent segments. Thus the current may be approximated by  
                 I        (   l   )              l   ^          (   l   )         ≈       ∑     n   =   1     N            I   n            Λ   n          (   l   )                l   ^     n          (   l   )                   (   9   )                               
 
         [0075]    in which the triangle function Λ n  about the n th  point on the segmented wire, as depicted in FIG. 5, is defined by  
                 Λ   n          (   l   )       =     {               l   -     l     n   -   1           Δ     n   -         ,     l   ∈     (       l     n   -   1       ,     l   n       )                         l     n   +   1       -   l       Δ     n   +         ,     l   ∈     (       l   n     ,     l     n   +   1         )                         (   10                               
 
         [0076]    where the unit vector Î n  is defined in terms of the unit vectors associated with the segments adjacent to the n th  point:  
                 l   ^     n     =     {                 l   ^       n   -       ,     l   ∈     (       l     n   -   1       ,     l   n       )                       l   ^       n   +       ,     l   ∈     (       l   n     ,     l     n   +   1         )               .               (   11   )                               
 
         [0077]    N is the number of basis functions and unknown current coefficients I n  in the finite series approximation (9) of the current. N unknowns are employed to represent the current on a wire having two free endpoints and modeled by N+1 straight-wire segments. In this traditional solution technique described here, N must be large enough to accurately model the geometric structure and vector direction of the current, even if a large number of unknowns is not required to approximate the current I(l) to the accuracy desired. The triangle basis functions overlap as suggested in FIG. 6 so an approximation with N terms incorporates, at most, N+1 vector directions of current on the wire. These point-by-point directions of current on a curved wire must be accounted for accurately by the N+1 unit vectors, yet N piecewise linear basis functions may be far more than may be needed to accurately represent the current I(l).  
         [0078]    Testing the integro-differential equation is accomplished by taking the inner product of (8) with the testing function  
                 Π   m          (   l   )       =     {           1   ,           l   ∈     (       l     m   -       ,     l     m   +         )                 0   ,           otherwise                     (   12   )                               
 
         [0079]    depicted in FIG. 7 for m=1, 2, . . . , N. The inner product of this testing pulse with a function of the variable l is defined by  
               〈     f   ,     Π   m       〉     =       ∫     l     m   -         l     m   +                f        (   l   )                 l     .                 (   13   )                               
 
         [0080]    Expanding the unknown current I with (9) and taking the inner product of (8) with (12) for m=1, 2, . . . , N yield a system of equations written in matrix form as 
         [ Z   mn   ][I   n   ]=[V   m ]  (14) 
         [0081]    where  
               Z   mn     =       -   j                     η     4      π                 k            {           k   2     2          [         (         Δ     m   -                l   ^       m   -       ·       l   ^       n   -           +       Δ     m   +                l   ^       m   +       ·       l   ^       n   -             )            ∫     l     n   -   1         l   n                Λ   n          (     l   ′     )            K        (     R   m     )                 l   ′             +       (         Δ     m   -                l   ^       m   -       ·       l   ^       n   +           +       Δ     m   +                l   ^       m   +       ·       l   ^       n   +             )            ∫     l   n       l     n   +   1                  Λ   n          (     l   ′     )            K        (     R   m     )                 l   ′               ]       +       1     Δ     n   -                ∫     l     n   -   1         l   n              K        (     R     m   +       )                 l   ′             -       1     Δ     n   +                ∫     l   n       l     n   +   1                K        (     R     m   +       )                 l   ′             -       1     Δ     n   -                ∫     l     n   -   1         l   n              K        (     R     m   -       )                 l   ′             +       1     Δ     n   +                ∫     l   n       l     n   +   1                K        (     R     m   -       )                 l   ′               }               (   15   )                               
 
         [0082]    is an element of the N×N impedance matrix with  
               R   m     =     {                   4        a   2          sin   2            φ   ′     2       +       (       l   m     -     l   ′       )     2         ,             l   m                     and                     l   ′                     on  same  segment                                r   m     -     r   ′            2     +     a   2         ,           otherwise                
          and                 (   16   )                 R     m   ±       =     {                 4        a   2          sin   2            φ   ′     2       +       (       l     m   ±       -     l   ′       )     2         ,             l     m   ±                       and                     l   ′                     on  same  segment                                r     m   ±       -     r   ′            2     +     a   2         ,             otherwise     .                     (   17   )                               
 
         [0083]    When the source (r′ or l′) and observation (r or l=(l m± ,l m )) points reside on the same straight wire segment of radius a, as in FIG. 8 the exact kernel given by  
               K        (     l   ,     l   ′       )       =       1     2      π              ∫     -   π     π                     -   j                   k                 R       R               φ   ′                     (   18   )                               
 
         [0084]    is used. Otherwise for source and observation points on different straight-wire segments (cf FIG. 9), the exact kernel is approximated by the so-called reduced kernel,  
               K        (     l   ,     l   ′       )       =                -   j                   k                 R       R     .             (   19   )                               
 
         [0085]    The approximation below, which is excellent when the segment lengths are small compared with the wavelength, is employed in arriving at the first two terms of (15):  
                 〈           l   ^          (   l   )       ·     f        (   l   )         ,       Π   m          (   l   )         〉     ≈       f        (     l   m     )       ·       [         1   2          Δ     m   -              l   ^       m   -         +       1   2          Δ     m   +              l   ^       m   +           ]     .                            (   20   )                               
 
         [0086]    The same approximation can be used to compute the elements of the excitation column vector,  
                 V   m     =       〈         -       E   i          (   l   )         ·     l   ^       ,     Π   m       〉     ≈       -       E   i          (     l   m     )         ·     [         1   2          Δ     m   -              l   ^       m   -         +       1   2          Δ     m   +              l   ^       m   +           ]           ,           (   21   )                               
 
         [0087]    where E i (l m ) is the known incident electric field at point l m  on the wire. Of course, if desired the left hand side of (21) can be evaluated numerically in those situations in which the incident field varies appreciably over a subdomain. We also point out that testing with pulses allows one to integrate directly the second term on the left side of (8). The derivative of the piecewise linear current in (8) leads to a pulse doublet (for charge) over two adjacent straight wire segments. These operations on the second term in the left side of (8) lead to the last four integrals in (15).  
         [0088]    IV. Improved Solution Technique  
         [0089]    In this section a new technique for solving the curved-wire integral equation is presented. It is very efficient for tortuous wires on which the actual variation of the current is modest, a situation which often occurs when the length of wire in a given curve is small relative to wavelength, regardless of the degree of curvature. Composite basis and testing functions are introduced as an extension of the functions of the traditional solution method outlined in Section III. The composite basis function serves to uncouple the number of straight segments needed to model the curved-wire geometry and the vector direction of the current from the number of unknowns needed to accurately represent the current on the wire. This new basis function is a linear combination of appropriately weighted generic basis functions, e.g., basis functions (9) in the traditional method outlined in Section III, and is defined over a number of contiguous straight segments. This new basis function is referred to as a composite basis function since it is constructed from others. Even though the solution method can incorporate any number of different generic basis and testing functions, the piecewise linear or triangle basis function and the pulse testing function are adopted here to facilitate explanation. Also, this pair leads to a very efficient and practicable solution scheme.  
         [0090]    The notion of a composite triangle made up of constituent triangles is suggested in FIG. 10. For simplicity in illustration, the composite triangle is shown over a straight line though in practice it would be over a polygonal line comprising straight-line segments, which approximate the curved wire axis. The q th  composite vector triangle function can be constructed as  
                     Λ   ~     q          (   l   )              l   ^     q       =       ∑     i   =   1       N   q              h   i   q            Λ   i   q          (   l   )              l   ^     i   q                 (   22   )                               
 
         [0091]    in which Λ i   q  is the i th  constituent triangle defined by  
                   Λ   i   q          (   l   )              l   ^     i   q       =     {                 l   -     l   i   q         Δ     i   -     q              l   ^       i   -     q       ,     l   ∈     (       l     i   -   1     q     ,     l   i   q       )                           l   i   q     -   l       Δ     i   +     q              l   ^       i   +     q       ,     l   ∈     (       l   i   q     ,     l     i   +   1     q       )                         (   23   )                               
 
         [0092]    and illustrated in FIG. 11. When q is used as a superscript it identifies a parameter related to the q th  composite triangle function. The constitutive elements of the q th  composite basis function are denoted by the subscript i. The parameter h i   q  is the weight or magnitude of the i th  constituent triangle within {tilde over (Λ)} q . These weights are functions of the segment lengths within each composite basis function and are adjusted so that the ordinate to the composite triangle is a linear function of displacement along the polygonal line which forms the base of the composite triangle. For example, for five constituent triangles in the q th  composite triangle of FIG. 10, the weights h 1   q , and h 2   q  are  
               h   1   q     =       Δ   1   q         Δ   1   q     +     Δ   2   q     +     Δ   3   q                 (   24   )                 h   2   q     =           Δ   1   q     +     Δ   2   q           Δ   1   q     +     Δ   2   q     +     Δ   3   q         .             (   25   )                               
 
         [0093]    The other weights are computed in a similar fashion. The parameter N q  is the number of triangle functions Λ i   q  employed to represent {tilde over (Λ)} q . The example composite basis function of FIG. 10 is illustrated as the sum of five identical constituent triangles, but, of course, the constituents need not be the same if convenience or efficiency dictates otherwise. Also, this composite basis function is illustrated without the vector directions associated with each subdomain. In general the individual straight-wire segments over which a composite basis function is defined may each have a different vector direction. Finally, the current expanded with a reduced number of unknowns Ñ is  
                 I        (   l   )              l   ^          (   l   )         =       ∑     q   =   1       N   ~                I   ~     q              Λ   ~     q          (   l   )                l   ^     q          (   l   )                   (   26   )                               
 
         [0094]    where {tilde over (Λ)} q (l)Ĩ q (l) is the q th  vector composite basis function defined earlier in (22) and Ĩ q  is its unknown current coefficient. It is worth noting that constituent triangles are employed above to construct composite triangles but, if desired, they could be used to construct other basis functions, e.g., an approximate, composite piecewise sinusoidal function.  
         [0095]    If the number of unknowns in a solution procedure is reduced, then, of course, the number of equations must be reduced too which means that the testing procedure must be modified to achieve fewer equations. This is easily accomplished by defining composite testing pulses, compatible with the composite basis functions, as a linear combination of appropriately weighted constituent pulses. An example composite test pulse is depicted in FIG. 12. Such a p th  composite testing pulse is defined by  
                   Π   ~     p          (   l   )       =       ∑     k   =   1       N   p              u   k   p            Π   k   p          (   l   )                   (   27   )                               
 
         [0096]    where the constituent pulses associated with this p th  pulse are  
                 Π   k          (   l   )       =     {           1   ,           l   ∈     (       l     k   -     p     ,     l     k   +     p       )                 0   ,           otherwise                     (   28   )                               
 
         [0097]    and shown in FIG. 13. If with every constituent triangle there were associated a corresponding constituent pulse, then the testing functions {tilde over (π)} p  would overlap, which is not desired and can be avoided by selecting the weight u k   p  to be 0 or 1 depending upon whether or not the k th  constituent pulse in {tilde over (π)} p  is to be retained. To this end, the inner product of (13) is modified in the composite testing procedure to become  
               〈     f   ,       Π   ~     p       〉     =       ∑     k   =   1       N   p              u   k   p            ∫     l     k   -     p       l     k   +     p              f        (   l   )                 l     .                     (   29   )                               
 
         [0098]    Now that we have described the new basis and testing functions, we substitute the current expansion of (26) into (8) and form the inner product (29) of the resulting expression with {tilde over (π)} p  for p=1, 2, . . . , Ñ. This yields the following matrix equation having a reduced number (Ñ) of unknowns and equations: 
         [ {tilde over (Z)}   pq   ][Ĩ   q   ]=[{tilde over (V)}   p ]  (30) 
         [0099]    where  
                 Z   ~     pq     =       ∑     k   =   1       N   p              u   k   p            ∑     i   =   1       N   q              h   i   q          {       -   j                       η     4      π                 k       [         k   2     2          {         (         Δ     k   -     p              l   ^       k   -     p     ·       l   ^       i   -     q         +       Δ     k   +     p              l   ^       k   +     p     ·       l   ^       i   -     q           )            ∫     l     i   -   1     q       l   i   q                Λ   i   q          (     l   ′     )          K                  (     R   k   p     )               l   ′         +       (         Δ     k   -     p              l   ^       k   -     p     ·       l   ^       i   +     q         +       Δ     k   +     p              l   ^       k   +     p     ·       l   ^       i   +     q           )            ∫     l   i   q       l     i   +   1     q                Λ   i   q          (     l   ′     )            K        (     R   k   p     )                 l   ′               }           +       1     Δ     i   -     q              ∫     l     i   -   1     q       l   i   q              K        (     R     k   +     p     )                 l   ′             -       1     Δ     i   +     q              ∫     l   i   q       l     i   +   1     q              K        (     R     k   +     p     )                 l   ′             -       1     Δ     i   -     q              ∫     l     i   -   1     q       l   i   q              K        (     R     k   -     p     )                 l   ′             +       1     Δ     i   +     q              ∫     l   i   q       l     i   +   1     q              K        (     R     k   -     p     )                 l   ′               ]       }                         (   31   )                               
 
         [0100]    represents an element of the reduced-rank (Ñ×Ñ) impedance matrix. At this point the reader is cautioned to distinguish between the index k which only appears in (31) as a subscript and the wave number k=ω{square root}{square root over (με)}. The distances R k±   p  and R k   p  are given in (16) or (17) with m replaced by index k, and the forcing function is given by  
           V   ~     pq     =     -       ∑     k   =   1       N   p              u   k   p            ∫     l     k   -     p       l     k   +     p                  E   i          (     l   k   p     )       ·       l   ^          (   l   )                   l     .                                     
 
         [0101]    One could compute the terms within the reduced-rank impedance matrix directly from (31). However, this would require more computation time than needed to fill the original impedance matrix of (14) since some constituent triangles within adjacent composite basis functions have the same support (FIG. 14). The constituent triangles within the overlapping portions of two adjacent composite basis functions differ only in the weight h l   q . Therefore (31) incorporates redundancies which should be avoided. Also, a study of (15) and (31) reveals that the term within the braces of (31) is identical to Z mn  of (15) if subscript i is replaced by n, subscript k by m, and the superscripts p and q are suppressed. Hence, the elements {tilde over (Z)} pq  of the reduced-rank matrix can be computed from the elements Z mn  of the original matrix by means of the transformation  
                 Z   ~     pq     =       ∑     k   =   1       N   p              u   k   p            ∑     i   =   1       N   q              h   i   q          Z   ki   pq                     (   32   )                               
 
         [0102]    where Z ki   pq  is a term in the original impedance matrix Z mn  of (15). The key to selecting appropriate Z mn  term is the combination of indices p, q, k, and i. The index p (q) indicates a group of rows (columns) in [Z mn ] which are ultimately combined by the transformation in (32) to form the new matrix. The appropriate matrix element Z kl   pq  in [Z mn ] is determined by intersecting the k th  row within the set of rows identified by index p with the i th  column of the group of columns specified by index q. Of course the groupings of rows and columns are determined when one defines the composite basis and testing functions.  
         [0103]    A transformation for computing the reduced-rank matrix [{tilde over (Z)} pq ] from the traditional matrix [Z mn ] which is more efficient than is the construction of the matrix from (31) can be developed. The key to this transformation is (32). First, two auxiliary matrices [L pm ] and [R nq ] are constructed and, then, the desired transformation is expressed as 
         [ {tilde over (Z)}   pq   ]=[L   pm   ][Z   mn   ][R   nq ]  (33) 
         [0104]    where  
                 [     L   pm     ]     =     [             u   1   1          u   2   1                   …                   u     N   1     1                                               ⋯                   0                           u   1   2          u   2   2                   …                   u     N   2     2                                                                                                       u   1   3          u   2   3                   …                   u     N   3     3                                               ⋮           ⋮                               ⋰                                                                                                   u   1   p          u   2   p                   …                   u     N   p     p                                                                                                   ⋰                       0                   ⋯                                               u   1     N   ~            u   2     N   ~                     …                   u     N     N   ~         N   ~               ]            
          and             (   34   )                 [     R   nq     ]     =     [                      h   1   1                                                         ⋯       0             h   2   1                                                                     ⋮           ⋮         h   1   2                                                                           h     N   1     1           h   2   2                                                                                     ⋮         h   1   3                                                                           h     N   2     2           h   2   3                                                                                     ⋮                                                                                     h     N   3     3                                                                                                 ⋰                                                                                                 h   1   q                                                                                       h   2   q                                                                                     ⋮                                                                                     h     N   q     q                                                                                                 ⋰                                                                                                 h   1     N   ~                                                                                         h   2     N   ~               ⋮                                                                   ⋮           0       ⋯                                                         h     N     N   ~         N   ~                        ]             (   35   )                               
 
         [0105]    It is easy to show that the above matrix transformation is equivalent to (32).  
         [0106]    An alternative development of the transformation, which renders the meaning and construction of the matrices [L pm ] and [R nq ] more transparent is presented. We begin with the traditional N×N system matrix equation, 
         [ Z   mn   ][I   n   ]=[V   m ]  (36) 
         [0107]    which is to be transformed to the Ñ×Ñ reduced-rank matrix equation 
         [ {tilde over (Z)}   pq   ][Ĩ   q   ]=[{tilde over (V)}   p ]  (37) 
         [0108]    The number of unknown current coefficients in the original system of equations (36) is reduced by expressing the Ñ coefficients Ĩ q  as linear combinations of the N coefficients I n  (Ñ&lt;N). The Ĩ q  are constructed from the I n  by means of a scheme which accounts for the representation of the composite basis functions in terms of the original triangles on the structure. The resulting relationships among the original and the composite coefficients are expressed as 
         [ I   n   ]=[R   nq   ][Ĩ   q ]  (38) 
         [0109]    where [R nq ] embodies weights of the constituent triangles needed to synthesize composite basis function triangles. The matrix [R nq ] directly combines unknown current coefficients consistent with the composite basis functions to result in a reduced number of unknowns. The construction is simple. If the triangle n from the original basis functions is to be used in the q th  composite basis function, the appropriate weight of this triangle is placed in row n and column q of [R nq ]. Otherwise zero is placed in this position. Again we point out that a given triangle may appear in more than one composite basis function. After substituting (38) into (36) we arrive at a modified system of linear equations 
         [ Z   mn   ][R   nq   ][Ĩ   q   ]=[V   m ]  (39) 
         [0110]    which has a reduced number (Ñ) of unknowns but the original number (N) of equations. To reduce the number of equations to Ñ, tested linear equations are selectively added, which is accomplished by pre-multiplying (39) by [L pm ] to arrive at 
         [ L   pm   ][Z   mn   ][R   nq   ][Ĩ   q   ]=[L   pm   ][V   m ]  (40) 
         [0111]    The identifications, 
         [ {tilde over (Z)}   pq   ]=[L   pm   ][Z   mn   ][R   nq ]  (41) 
         [0112]    and 
         [ {tilde over (Z)}   p   ]=[L   pm   ][V   m ],  (42) 
         [0113]    in (40) lead to the desired expression (37). The matrix [L pm ] effectively creates composite testing functions from the original testing pulses. If the p th  composite testing pulse contains the m th  testing pulse from the original formulation, a one is placed in row p and column m of [L pm ]. Otherwise, a zero is placed in this position.  
         [0114]    There are other important considerations in the implementation of this technique. Again, we label the number of basis functions in the traditional formulation N and the number of composite basis functions Ñ. In the previous section the number of constituent triangles for the q th  composite basis function is designated N q . Here for ease of implementation it is convenient to chose N q  to be the same value for every q, which we designate τ (N q =τ for all q ). Also, in the present discussion, we restrict τ to be one of the members of the arithmetic progression 5, 9, 13, 17 . . . ,. With τ one of these integers, half-width constituent pulses are not required within the composite testing functions. N must be sufficiently large to ensure accurate modeling of the wire geometry and vector direction of the current as well as to preserve the numerical accuracy of the approximations. In addition, Ñ must be large enough to accurately represent the variation of the current. A convergence test must be conducted to arrive at acceptable values of N and Ñ. Also, N,Ñ and τ must be defined carefully so that a value of τ in the arithmetic progression will allow an N×N matrix to be reduced to an Ñ×Ñ matrix. The following formula is useful for determining relationships between N and Ñ, for a given value of τ, in the case of a general three-dimensional curved wire (without junctions):  
               N   ~     =       2                 N     +     1     τ   +   1       -   1.             (   43   )                               
 
         [0115]    For a wire structure with a junction, e.g., a circular loop, where overlapping basis functions typically are used in the traditional formulation to satisfy Kirchhoff&#39;s current law, (43) becomes  
               N   ~     =         2      N       τ   +   1       .             (   44   )                               
 
         [0116]    Once N, Ñ and τ are determined, it is easy to write a routine which determines the original basis and testing functions to be included in the composite functions. This information is then stored in the matrices [L pm ] and [R nq ].  
         [0117]    In the above, composite triangle expansion functions are synthesized from generic triangle functions but one could as well, if desired, approximate other composite expansion functions, e.g., “sine triangles” by adjustment of the coefficients h i   q . Similarly, other approximate testing functions could be created by adjustment of the factors u k   p . Thus, a reduced-rank solution method with composite expansion and testing functions different from triangles and pulses could be readily created from the techniques discussed in this section. Only h i   q  and u k   p , peculiar to the functions selected in the method to be implemented, must be changed in (32) in order to arrive at the appropriate reduced-rank matrix elements {tilde over (Z)}pq. If [L pm ] of (34) were replaced by [R nq ] T  in (33) where [R nq ] is defined in (35) and T denotes transpose, then the resulting reduced-rank matrix [{tilde over (Z)} pq ] would be that for a method which employs composite triangle expansion and (approximate) composite triangle testing functions.  
         [0118]    V. Results  
         [0119]    Results obtained by solving the integral equation of (15) with the improved solution method developed above are presented in this section as are values of current determined by the traditional method. In some cases data obtained from the literature are displayed for comparison. Results are presented for the wire loop, an Archimedian spiral antenna, and several different helical antennas and scatterers.  
         [0120]    Current values on a small wire loop antenna are depicted in FIG. 15. The loop is modeled by 32 linear segments (and 32 unknowns) in the traditional solution technique. Also shown are values obtained from the new solution method with eight composite basis functions (eight unknowns) each having five constituent triangles constructed on twenty four linear segments. These current values compare well with those from the traditional solution and with data from [2] where the loop is modeled with eight unknowns on quadratic segments. There is slight disagreement at the driving point which is to be expected (with eight unknowns) near a delta gap source where the current varies markedly. To investigate this discrepancy we use three triangle basis functions in the vicinity of the delta-gap source and do not form composite triangles in this region. The results are shown in FIG. 16. Here the loop problem has been solved with 28 unknowns for the traditional method and twelve unknowns for the composite basis function solution. It is seen that the agreement is excellent even in the vicinity of the delta-gap source.  
         [0121]    The improved solution method is applied to a four arm Archimedian spiral antenna. This antenna is chosen since it is used in [2] to illustrate the usefulness of the quadratic subdomains for wires having significant curvature. A description of the geometry of Archimedian spiral antennas is found in [17] and [18]. The antenna is excited by a delta gap source on each arm located near the junction of the four arms. The results presented in this section are for mode 2 excitation [19]. The antenna is also modeled by the traditional technique with 725 unknowns on each arm (725*4+3=2903). In [17] the authors implement a discrete body of revolution technique so that the number of unknowns needed for one arm is sufficient for solving the problem. Since our goal is to employ the data of [17] to demonstrate the accuracy of our method and not to create the best analytical tool for the Archimedian spiral antenna, we solve this problem by including the same number of linear segments on each arm and placing overlapping triangles at the wire junction to enforce Kirchhoff&#39;s current law. In [2] it is found that each arm requires 504 linear segments to obtain an accurate solution. They also obtain accurate values of the current with 242 quadratic segments. We reproduce these results with our improved solution method as illustrated in FIG. 17-FIG. 19. The number of unknowns for each arm is 725 for the traditional technique and 241 for the improved method. In each composite basis function there are five constituent triangles. In FIG. 17 the difference in the solution of the current for the two methods is seen to be negligible. Good agreement is also achieved for the current magnitude (cf. FIG. 18). A favorable comparison with data from [2] is observed in FIG. 19. Since the symmetry in the geometry is not used to further reduce the number of unknowns required for the structure, the actual number of unknowns in the impedance matrices are 2903 and 967, respectively. The computation times for the various routines of the FORTRAN 90 code are presented in the table below. All times are for runs on a 375 MHz DEC Alpha processor. The time study shows that the reduction technique is successful in significantly reducing matrix solve time for this four-arm Archimedian spiral antenna. A standard linear equation solution method is employed to solve both sets of linear equations since the objective of this comparison is to delineate the enhanced efficiency of the reduced-rank method.  
                                           TABLE I                           COMPUTATION TIMES FOR ARCHIMEDIAN SPIRAL                Event   Time in Seconds                            Fill matrix N = 2903   1020           Solve matrix equation N = 2903   1329           Reduce matrix from 2903 to 967   5.54           Solve reduced matrix equation N = 967   45.81           Traditional method total time   2349           Improved method total time   1071                      
 
         [0122]    Consider next a ten-turn helix having a total wire length of 0.5λ and illuminated by a plane wave. The geometry of the helical scatterer is depicted in FIG. 20. The current shown in FIG. 21 is “converged” when the number of unknowns in the traditional solution technique reaches 259. Thus one concludes that 260 linear segments are required to accurately represent the geometry of this structure and vector nature of the current. We determine convergence by examining the real and imaginary parts of the current along the structure. When changes in the current are sufficiently small as the number of segments is increased, convergence is assumed [2]. The results of a convergence test show that an accurate solution of the current can be achieved with 51 composite basis functions. The number of constituent triangles in each basis function in this case is nine. We note that the solution with 27 composite basis functions differs only slightly from the converged solution.  
         [0123]    The current is shown in FIG. 22 for another helical scatterer of geometry similar to that described above and subject to the same excitation and geometry similar to that described above. The circumference of each turn of this ten-turn helix is 0.035λ making the total wire length 0.35λ. These results are given as an example to illustrate that the composite basis function scheme works well with curved-wire structures having a wire length which is not an integer multiple of half wavelength.  
         [0124]    The data of FIG. 23 are for a 50-turn helix having a total wire length of 2λ and illuminated by a plane wave traveling in the positive x direction. One sees that 27 unknowns are adequate to accurately represent the current along the helix. However, 1483 unknowns are required in the traditional solution method since many linear segments are required to define the 50-turn structure and the vector properties of the current. In this example there are 105 constituent triangles in each composite basis function. The table below shows the computational savings enjoyed by the method of this paper.  
                                           TABLE II                           COMPUTATION TIMES FOR FIFTY-TURN       HELIX                Event   Time in Seconds                            Fill matrix N = 1483   300           Solve matrix equation N = 1483   267           Reduce matrix rank from 1483 to 27   1.84           Solve reduced matrix equation N = 27   Negligible           Traditional method total time   567           Improved method total time   302                      
 
         [0125]    Next we illustrate the prowess of the solution technique for helical antennas. Specifically the data presented in FIG. 24 and FIG. 25 are for helical antennas driven above a ground plane by a delta gap source. The geometry of the helix is given in FIG. 20 and the ground plane is located at z=0. The data of the improved method compare well with those of the traditional solution technique, but, again, there is a slight difference in the currents at the ground plane due to the nature of the delta gap source. In each of these figures the number of unknowns given is the number for the structure plus its image, but data are plotted only for the part of the structure above the ground plane. Since there are many turns, the number of segments needed to represent the geometry of the antenna and its image is large. The number of unknowns is reduced from N=917 in the traditional method to N=53 in the improved technique. Of course, one could employ image theory to modify the integral equation which could be solved by the new method with an even more dramatic savings in computer resources.  
         [0126]    The last example is a five-turn helical antenna over an infinite ground plane, driven by a delta gap source. This structure is included here because it is used in [6] to exhibit the accuracy of a technique employing basis and testing functions defined over arcs of constant radii. It is modeled by straight wire segments in [20]. In [6] the authors discretize the antenna into fifteen arcs and then compare solutions of 135 unknowns with forty-five unknowns. They find that forty-five unknowns is enough to obtain an accurate solution for the current when the geometry is defined by arcs. We reproduce these results except that the antenna geometry is defined by many straight wire segments. In the method of this paper we include the unknowns on the image (269 unknowns on the antenna plus its image corresponds to 135 unknowns on the antenna above the ground plane). Likewise, 89 unknowns on the antenna and image are equivalent to 45 unknowns on the antenna. We find that helical antennas require a minimum of 25 unknowns per turn in the traditional solution technique in order to represent the geometry. In order to reduce the number of unknowns over the antenna and its image from 269 to 89, each composite basis function is constructed with 5 constituent triangles. A qualitative comparison of our data and that of [6] suggests agreement in the two methods.  
         [0127]    VI. Conclusions  
         [0128]    The solution method presented in this paper is very simple and practicable for reducing the rank of the impedance matrix for curved-wire structures. It should be mentioned that rank reduction is realized only when the number of segments needed to model the geometry and vector direction of the current exceeds the number of unknown current coefficients necessary to characterize the variation of the current. We define composite basis and testing functions as the sum of constituents over linear segments on a wire and arrive at a new impedance matrix of reduced rank. It is shown how this reduced-rank matrix can be determined from the original impedance matrix by a matrix transformation. Thus one advantage of this technique is that it can be applied to almost any existing curved-wire codes which define basis and testing functions over straight-wire segments or curved-wire segments.  
         [0129]    Dramatic savings in matrix solve time are realized for the cases of the four-arm Archimedian spiral antenna and the helical antenna. The benefits for reducing unknowns on, for example, a helical antenna become much more significant as the number of turns increases. It should be pointed out that this method does not reduce matrix fill time since the elements of the original impedance matrix are computed as a step in the determination the elements of the reduced-rank matrix. Problems involving large curved-wire structures can be solved readily by this method, e.g., a straight wire antenna loaded with multiple, tightly wound helical coils and an array of Archimedian spiral antennas. The principles described here can be used in addition to other methods such as those based upon iteration.  
       REFERENCES  
       [0130]    [1] R. F. Harrington,  Field Computation by Moment Methods.  Malabar, FL: Krieger, 1968.  
         [0131]    [2] N. J. Champagne, II, J. T. Williams, and D. R. Wilton, “The use of curved segments for numerically modeling thin wire antennas and scatterers,”  IEEE Trans. Antennas Propagat.,  vol. 40, pp. 682-689, June 1992.  
         [0132]    [3] E. K. N. Yung and R. S. K. Wong, “Analysis of an array of circular loops.”  Annals of Telecommunications,  vol. 48, no. 9-10, pp. 491-497, 1993.  
         [0133]    [4] E. K. N. Yung and R. S. K. Wong, “Analysis of a thin wire circular loop antenna,”  International Journal of Numerical Modelling: Electronic Networks, Devices and Fields,  vol. 7, pp. 189-199, 1994.  
         [0134]    [5] M. A. Jensen and Y. Rahmat-Sanii, “Electromagnetic characteristics of superquadric wire loop antennas,”  IEEE Trans. Antennas Propagat.,  vol. 42, pp. 264-269, February 1997.  
         [0135]    [6] E. K. N. Yung and R. S. K. Wong, “Analysis of a wire antenna of arbitrary shape,” J.  Electromagnetic Waves and Applications,  vol. 9, no. 7/8, pp. 855-869, 1995.  
         [0136]    [7] S. K. Khamas, G. G. Cook, and R. J. Waldron, “Moment-Method analysis of printed circular wire-loop antenna using curved piecewise sinusoidal subdomain basis and test functions,”  IEEE Trans. Antennas Propagat.,  vol. 44, pp. 1303-1305, September 1996.  
         [0137]    [8] S. K. Khamas, G. G. Cook, “Moment-Method analysis of printed wire spirals using curved piecewise sinusoidal subdomain basis and test functions,”  IEEE Trans. Antennas Propagat.,  vol. 45, pp. 1016-1022, June 1997.  
         [0138]    [9] G. G. Cook and S. K. Khamas, “Efficient moment method for analysing printed wire loop antennas,”  IEE Proc. - Microw. Antennas Propag.,  vol. 144, no.5, pp.364-366, October 1997.  
         [0139]    [10] S. K. Khamas, et. al., “Moment method analysis of printed single-arm wire spiral antennas using curved segrnents,”  IEE Proc. - Microw. Antennas Propag.,  vol. 144, no. 4., pp. 261-265, August 1997.  
         [0140]    [11] B. M. Kolundzija and B. D. Popovic, “Entire domain Galerkin method for analysis of generalised wire antennas and scatterers,”  IEE Proc. - Microw. Antennas Propag.,  vol. 139, no. 1 pp. 17-24, February 1992.  
         [0141]    [12] S. D. Rogers and C. M. Butler, “Reduced Rank Matrices for Curved Wire Structures,” Digest of IEEE APS/URSI Radio Science Meeting, July 1997.  
         [0142]    [13] S. D. Rogers, “Efficient Numerical Techniques for Curved Wires,” M.S.E.E. thesis, Clemson University, Clemson, S.C., 1997.  
         [0143]    [14] K. Kalbasi and K. R. Demarest, “A Multilevel Formulation of the Method of Moments,”  IEEE Trans. Antennas Propagat.,  vol. 41, pp. 589-599, May 1993.  
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         [0146]    [17] N. J. Champagne, II, “Method of moments formulations for thin wire antennas and scatterers using piecewise linear and curved wire segments,” M.S.E.E. thesis, Univ. Houston, Houston, Tex., 1991.  
         [0147]    [18] N. J. Champagne, II, J. T. Williams, R. M. Sharpe, S. U. Hwu, and D.R. Wilton, “Numerical modeling of impedance loaded multi-arm Archimedian spiral antennas,”  IEEE Trans. Antennas Propagat.,  vol. 40, pp. 102-108, January 1992.  
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         [0151]    [22] M. Ali, S. S. Stuchly, and K. Caputa, “Characteristics of bent wire antennas,”  J. Electromagnetic Waves and Applications,  vol. 9, no. 9, pp. 1149-1162, 1995.  
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         [0153]    The following is a detailed description of an exemplary genetic algorithm that can be used in accordance with the subject invention to obtain optimal antenna parameters for given design criteria. 
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
 
         [0154]    The following portion of this disclosure was created in Powerpoint for purposes of further describing the present invention. It particularly concerns bandwidth enhanced normal mode helical antennas. It begins by setting forth the objectives, considerations, and questions addressed in the beginning stages of development of the present invention. The affects of different physical antenna parameters on antenna performance are addressed by showing the affect on the VSWR by these variations.  
         [0155]    The remainder of the following disclosure portion shows several different antenna designs and in graphical form illustrates the respective performance of each. A straight wire antenna, a simple helix, and a triple helix are all examined. Each antenna is modified by the addition of various combinations of parasitic elements. The characteristics of each of these antennas are then illustrated. The VSWR, directivity, and input impedance are shown so that the different antennas having different combinations of parasitic elements can be analyzed effectively.  
         [0156]    This portion of the disclosure concludes by summarizing the results obtained from the different combinations. The conclusions drawn from these results are then set forth. It illustrates the initial indications that bandwidth improvements could be made by the addition of these parasitic elements. 
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
         
 
         [0157]    The following further portion of this disclosure was also created in Powerpoint for purposes of further describing the present invention. It particularly concerns the sleeve-cage monopole and sleeve helix for wide band operation. It sets forth the objectives, considerations, and questions addressed in the development of the present invention, and presents relevant background information and illustrations of the antennas discussed within.  
         [0158]    The VSWR, input impedance, and directivity are given for each antenna with and without the addition of the parasitic elements. Illustrations and graphical data for the cage monopole, the sleeve-cage monopole, the quadrifilar helix, and the sleeve-helix are presented. The relevant data for each is then shown in a table so that a side-by-side comparison can be made to clearly illustrate the improvements made in the antenna characteristics by the optimal placement of parasitic elements.  
         [0159]    The physical measurements and characterization values of various antennas optimized for various VSWR values are presented. This data is then presented in a comparison to several background antennas to illustrate the improvements in antenna performance made by the present invention.