Abstract:
A delayed decision feedback sequence estimation diversity receiver includes a section for extracting a plurality of reception signals by using a plurality of antennas when estimating a transmission signal from reception signals having undergone transmission path distortion, a section for combining impulse response sequences in transmission paths while canceling delayed wave components having the largest amplitudes in delayed wave component sequences in impulse response sequences in the respective transmission paths, and a section for performing signal estimation on the basis of a new impulse response sequence generated by combining the impulse response sequences. A delayed decision feedback sequence estimation method is also disclosed.

Description:
BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The present invention relates to a DDFSE (Delayed Decision Feedback Sequence Estimator) for estimating a transmission signal from a signal having undergone transmission path distortion caused by frequency selective fading due to a multipath effect in a radio channel in high-speed digital communication and, more particularly, to a delayed decision feedback sequence estimation diversity receiver which improves its signal estimation ability by combining antenna diversity with a DDFSE. 
   2. Description of the Prior Art 
   As a conventional apparatus designed to determine an optimal reception timing so as to estimate a transmission signal from reception signals having undergone transmission path distortion by using a DDFSE, the delayed decision feedback sequence estimation receiver disclosed in Japanese Unexamined Patent Publication No. 11-8573 is known. 
   The DDFSE is a signal estimator which has the merits of both an MLSE (Maximum Likelihood Sequence Estimator) having high signal estimation ability and a DFE (Decision Feedback Equalizer) with a small computation amount. 
     FIG. 1  is a block diagram showing the arrangement of a conventional DDFSE with a timing control function. 
   Assume that a reception signal  201  is a complex baseband signal expressed in a two-dimensional form. A transmission path estimator  202  is a block for obtaining the characteristics of a transmission path in the form of an impulse response. In general, the transmitting side sends a training signal before transmission of data, and the receiving side receives the training signal having undergone transmission path distortion, thereby obtaining transmission path characteristics. 
   An estimation region detector  203  performs a computation to find the timing at which the signal estimation ability is maximized. A DDFSE  204  performs signal estimation on the basis of the impulse response sequence obtained by the transmission path estimator  202  and the optimal timing obtained by the estimation region detector  203 . 
   If the impulse response sequence obtained by the transmission path estimator  202  has undergone transmission path distortion, it has a temporally wide waveform like the one shown in  FIG. 2 . In this case, this signal is expressed in the form of a discrete signal sampled at a symbol period T of the transmission signal.  FIG. 2  shows how the distortion spreads over a time  6 T (signal components a 2 , a 3 , and a 6  to a 10  are not shown because their amplitudes are regarded as 0). 
   Assume that the DDFSE with the timing control function is configured to perform transmission path estimation in 11 symbol periods. More specifically, the DDFSE performs signal estimation equivalent to an MLSE computation in the first three symbol periods, and cancels a component corresponding to the succeeding three symbol periods by a computation equivalent to a DFE computation. 
   The estimation region detector  203  can find an optimal timing by the following computation. 
   Let P be the power component used for signal estimation, which falls within a 3-symbol range (MLSE region), Q be the power component to be canceled, which falls within a 3-symbol range (DFE region), and R be the power in the remaining 5-symbol range (outside the estimation region). In this case, as P increases, the signal estimation ability increases. Q is irrelevant to the signal estimation ability because it is canceled. As R increases, the signal estimation ability decreases. As an evaluation function, we define:
 
 Z=P/R   (1)
 
   The signal estimation ability is maximized at the timing at which Z of equation (1) is maximized. 
   In general, an impulse response in a transmission path can be obtained accurately only within certain limits on the receiving side owing to the influences of noise and computation errors. For this reason, the signal component in the DFE region which should be completely canceled ideally is not completely canceled and left as a distortion component. This phenomenon becomes noticeable as the signal component in the MLSE region decreases and the signal component in the DFE region increases. 
   A decision feedback loop exists in the DDFSE. Once an error is made in signal estimation, therefore, the erroneous estimation result circulates within the loop, and a burst-like error called error propagation may occur. This error propagation is likely to occur as the component in the DFE region becomes large. In order to cope with this situation, the evaluation function expressed by equation (1) must be modified to determine the timing at which higher signal estimation ability can be obtained. To this end, we define an evaluation function given by:
 
 Z=P /( R+αQ )  (2)
 
   In equation (2), the coefficient α is a coefficient determined in accordance with the computation precision of an impulse response. 
   In the transmission path impulse response sequence shown in FIG.  2 ., the timings represented by:
 
 P =( a   0 ) 2 +( a   1 ) 2 +( a   2 ) 2   (3)
 
 Q =( a   3 ) 2 +( a   4 ) 2 +( a   5 ) 2   (4)
 
 R =( a   6 ) 2 +( a   1 ) 2 +( a   8 ) 2 +( a   9 ) 2 +( a   10 ) 2   (5)
 
are obtained as optimal timings for signal estimation by using either equation (1) or (2).
 
   If signal components that are received with delays are larger than other components as shown in  FIG. 3 , the timings obtained by equations (1) and (2) may differ from each other. In using equation (2), the timings are matched to delayed components that are received with delays by adjusting the coefficient a as per:
 
 P =( a   3 ) 2 +( a   4 ) 2 +( a   5 ) 2   (6)
 
 Q =( a   6 ) 2 +( a   7 ) 2 +( a   8 ) 2   (7)
 
 R =( a   9 ) 2 +( a   10 ) 2 +( a   0 ) 2 +( a   1 ) 2 +( a   2 ) 2   (8)
 
   This is because the estimation ability can be improved by performing signal estimation using a 4  and a 5  while regarding a 0  and a 1  as distortion components rather than by performing signal estimation using a 0  and a 1  with small amplitudes. 
     FIG. 4  shows the arrangement of this estimation region detector  203 . 
   A power calculator  701  obtains the power level of each symbol, which is the square value (the sum of the square value of a real part and the square value of an imaginary part) of each symbol, of the complex impulse response sequence output from the transmission path estimator  202 , and inputs the respective power levels to shift registers  702   a  to  702   j.    
   An adder  703  obtains a power value P of the signal component in the MLSE region. An adder  704  obtains a power value Q of the signal component in the DFE region. An adder  705  obtains a power value R of a signal component outside the estimation region for the DDFSE  204 . 
   Equations (3) and (6) are calculated by the adder  703 . Equations (4) and (7) are calculated by the adder  704 . Equations (5) and (8) are calculated by the adder  705 . 
   The power values P, Q, and R obtained by the adders  703 ,  704 , and  705  are used by an evaluation function calculator  706  to perform a computation based on equation (2). The evaluation function calculator  706  calculates equation (2) over 11 symbol periods, and detects the timing at which the value of Z is maximized. The evaluation function calculator  706  then outputs this timing to the DDFSE  204 . 
   In this manner, the DDFSE with the timing control function obtains the timing for signal estimation by using an evaluation function like equation (2), thereby obtaining an optimal timing for the DDFSE. 
   However, the following problem arises in the prior art described above. 
   In a transmission path impulse response sequence like the one shown in  FIG. 3 , if a 0  and a 1  are received in the MLSE region as optimal timings, a 4  and a 5  received in the DFE region are canceled by a 0  and a 1  having small amplitudes. At this time, if a slight error is included in a 0  or a 1 , the error is amplified when a 4  and a 5  are canceled, resulting in a deterioration in signal estimation ability. 
   If the values of a 4  and a 5  are large, the probability of occurrence of error propagation, i.e., continuous occurrence of errors upon occurrence of an error in signal estimation, increases. This also leads to a deterioration in signal estimation ability. 
   If a 4  and a 5  are received in the MLSE region, since a 0  and a 1  are received in neither the MLSE region nor the DFE region, these values are not effectively used for signal estimation and treated as distortions. This becomes a factor that degrades the signal estimation ability. That is, high signal estimation ability can be obtained by selecting neither of the former timing and the latter timing. 
   When a relatively large power component is set in the DFE region, as shown in  FIG. 2 , error propagation occurs more easily than when a large power component is not set in the DFE region. Therefore, a deterioration in signal estimation ability cannot be avoided. 
   SUMMARY OF THE INVENTION 
   The present invention has been made in consideration of the above situation in the prior art, and has as its object to provide a delayed decision feedback sequence estimation diversity receiver which can obtain high signal estimation ability. 
   In order to achieve the above object, according to the first aspect of the present invention, there is provided a delayed decision feedback sequence estimation diversity receiver characterized in that signals are received by two or more antennas, impulse response sequences in the respective transmission paths are obtained from the respective reception signals, components having the largest amplitude values among delayed wave components that are received with delays in these impulse response sequences are detected, and the impulse response sequences are combined so as to cancel the detected delayed wave components to generate a new impulse response sequence. 
   According to the second aspect of the present invention, there is provided a delayed decision feedback sequence estimation diversity receiver characterized in that signals are received by using two or more antennas, and the respective reception signals are combined so as to cancel components having the largest amplitude values among delayed wave components received with delays, thereby generating a new reception signal. 
   According to the third aspect of the present invention, there is provided a delayed decision feedback sequence estimation diversity receiver characterized in that signal estimation is performed by receiving a newly generated impulse response sequence and a newly generated reception signal and performing a computation for delayed decision feedback sequence estimation. 
   As is obvious from the respective aspects described above, according to the delayed decision feedback sequence estimation diversity receiver of the present invention, the overall power of delayed wave components is decreased by canceling components having the largest amplitudes among delayed wave components which cause a deterioration in signal estimation in a DDFSE by using a delayed wave canceler. 
   As a consequence, the power of delayed wave components which cause a deterioration in the DDFSE decreases, and hence the signal estimation ability of the DDFSE can be improved. 
   The above and many other objects, features and advantages of the present invention will become manifest to those skilled in the art upon making reference to the following detailed description and accompanying drawings in which preferred embodiments incorporating the principle of the present invention are shown by way of illustrative examples. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a block diagram showing the arrangement of a conventional delayed decision feedback sequence estimation diversity receiver having a timing control function; 
       FIGS. 2 and 3  are charts for explaining conventional impulse response sequences in transmission paths; 
       FIG. 4  is a block diagram showing the detailed arrangement of an estimation region detector in the prior art; 
       FIG. 5  is a block diagram showing a delayed decision feedback sequence estimation diversity receiver according to an embodiment of the present invention; 
       FIG. 6  is a block diagram showing the detailed arrangement of a delayed wave detector in the embodiment of the present invention in  FIG. 5 ; 
       FIG. 7  is a block diagram showing the detailed arrangement of a delayed wave canceler according to the embodiment of the present invention in  FIG. 5 ; and 
       FIGS. 8 and 9  are charts for explaining the impulse response sequences output from the delayed wave canceler according to the embodiment of the present invention in  FIG. 5 . 
   

   DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
   A preferred embodiment of the present invention will be described below with reference to the accompanying drawings. 
     FIG. 5  is a block diagram showing the arrangement of a delayed decision feedback sequence estimation diversity receiver according to an embodiment of the present invention. 
   Referring to  FIG. 5 , the delayed decision feedback sequence estimation diversity receiver includes transmission path estimators  103  and  104  for respectively obtaining transmission path complex impulse response sequences from complex baseband reception signals  101  and  102  input through input terminals T 1  and T 2  and received by two independent antennas. 
   The delayed decision feedback sequence estimation diversity receiver of the present invention includes delayed wave detectors  105  and  106  for detecting the positions and magnitudes of components having the largest amplitudes among delayed wave components from the complex impulse response sequences respectively obtained by the transmission path estimators  103  and  104 , a delayed wave canceler  107  for outputting an impulse response sequence obtained by canceling a component having the largest amplitude among delayed wave component sequences in the impulse response sequences output from the transmission path estimators  103  and  104  on the basis of the output signals from the delayed wave detectors  105  and  106 , and a delayed wave canceler  108  for outputting a complex baseband reception signal obtained by canceling a component having the largest amplitude among delayed wave components in the reception signals input through the input terminals T 1  and T 2 . 
   The delayed decision feedback sequence estimation diversity receiver also includes an estimation region detector  109  for determining an optimal timing for signal estimation from the impulse response sequence output from the delayed wave canceler  107 , and a DBFSE  110  for performing signal estimation by receiving the output signals from the delayed wave canceler  107 , estimation region detector  109 , and delayed wave canceler  108 . 
   The overall operation of this embodiment will be described next with reference to the arrangement shown in  FIG. 5 . 
   In this case, a 11-bit pseudo-random code is used as a training signal to allow the transmission path estimators  103  and  104  to obtain impulse response sequences based on multipath distortion in transmission paths during a 11-symbol period. As regions that can be estimated by the DDFSE (Delayed Decision feedback Sequence Estimator)  110 , a maximum likelihood sequence estimation region (MLSE region) and decision feedback equalization region (DFE region), each corresponding to three symbols, will be described below. 
   A transmission path is estimated on the transmitting side when a training signal is transmitted. The training signal generated from a 11-bit pseudo-random code on the transmitting side is input as the reception signal  101  through the input terminal T 1 . The transmission path estimator  103  obtains an impulse response sequence in the transmission path by performing a correlation computation between the reception signal  101  and a 11-bit pseudo-random code identical to that on the transmitting side. 
   As the 11-bit pseudo-random code, a Barker code (+1, +1, +1, −1, −1, −1, +1, −1, −1, +1, −1) is used, and the received training signal is represented by r(n). In this case, an output signal h(n) from the transmission path estimator  103  is given by
 
 h ( n )=r( n− 10)+ r ( n− 9)+ r ( n− 8)− r ( n− 7)− r ( n− 6)− r ( n− 5)+ r ( n− 4)− r ( n− 3)− r ( n− 2)+ r ( n− 1)− r ( n )  (9)
 
where n is an integer having a symbol period.
 
   This output signal h(n) becomes an impulse response sequence in the transmission path. Since a baseband reception signal is generally a two-dimensional signal, the signal given by equation (9) is also a two-dimensional signal. 
   The transmission path estimator  104  receives the reception signal through the input terminal T 2 , which is received by using an antenna different from that used for the reception signal  101 , and performs a correlation computation with a 11-bit pseudo-random code in the same manner as described above, thereby obtaining an impulse response sequence in the transmission path. 
   Assume that the impulse response sequence obtained by the transmission path estimator  103  from the reception signal is the sequence shown in  FIG. 2 , and the impulse response sequence obtained by the transmission path estimator  104  from the reception signal  102  is the sequence shown in  FIG. 3 . 
   The delayed wave detector  105  detects the timing, real component, and imaginary component of a 4  in  FIG. 2  which is the component having the largest amplitude in the delayed wave sequence. In this case, the timing is represented by m 1 , and the component is represented by p 1 +j×q 1 . Note that j is an imaginary unit. 
     FIG. 6  shows an example of the arrangement of the delayed wave detector  105 . 
   The two-dimensional impulse response sequence value input from the transmission path estimator  103  is shifted at a symbol cycle by using shift registers  801   a  to  801   e.    
   The magnitudes of impulse responses at three symbols, i.e., the fourth to sixth symbols, of the signal input from the transmission path estimator  103  are compared with each other. 
   The impulse response value at the fourth symbol is output from the shift register  801   c , and its power level is obtained by a power calculator  802 . The impulse response value at the fifth symbol is output from the shift register  801   d , and its power level is obtained by a power calculator  803 . The impulse response value at the sixth symbol is output from the shift register  801   e , and its power level is obtained by a power calculator  804 . 
   The power levels at the fourth, fifth, and sixth symbols, respectively obtained by the power calculators  802 ,  803 , and  804 , are compared by a comparator  805  to determine a specific symbol at which the highest level is obtained. The corresponding information (timing m 1 ) is output to a selector  806 . The selector  806  outputs the component (p 1 +j×q 1 ) having the largest amplitude among the components at the fourth, fifth, and sixth symbols in the impulse response sequence. 
   The other delayed wave detector  106  has the same arrangement as that of the delayed wave detector  105 . The delayed wave detector  106  obtains the timing, real component, imaginary component of a 4  in  FIG. 3 . In this case, the timing is represented by m 2 , and the component is expressed by p 2 +j×q 2  as a complex number. 
   The delayed wave canceler  107  generates an impulse response sequence by canceling the largest component of a delayed wave using the output signals from the delayed wave detectors  105  and  106 . This computation is performed as follows. 
   The impulse response sequence output from the transmission path estimator  103  is represented by h 1 ( k ), and the impulse response sequence output from the transmission path estimator  104  is represented by h 2 ( k ). In this case, k represents the timing of symbol periods and takes an integer from 0 to 10. Letting dm be the difference between a timing m 1  and a timing m 2 , the computation by the delayed wave canceler  107  is expressed as
 
 h   1 ( k )×( p   2 + j×q   2 )− h   2 ( k−dm )×( p   1 + j×q   1 )  (10)
 
In mathematical expression (10), (k−dm) is the remainder of 11.
 
   The computation result on mathematical expression (10) becomes a new impulse response sequence.  FIG. 8  shows such a case. 
   When the component having the largest amplitude among delayed wave components is canceled, the ratio of a delayed component to a corresponding direct wave component increases, and high signal estimation ability can be obtained. 
     FIG. 7  shows an example of the arrangement of the delayed wave canceler  107 . 
   A computation based on mathematical expression (10) can be performed by using a complex multiplier  901 , complex multiplier  902 , and complex subtractor  903 , and an impulse response sequence obtained by canceling the delayed wave component having the largest amplitude can be output. 
   The new impulse response sequence is obtained by the delayed wave canceler  107 . A new reception signal must be obtained accordingly. Letting S 1 ( k ) be the reception signal  101 , and S 2 ( k ) be the reception signal  102 , the output signal from the delayed wave canceler  108  is given by
 
 S   1 ( k )×( p   2 + j×q   2 )− S   2 ( k−dm )×( p   1 + j×q   1 )  (11)
 
   The delayed wave canceler  108  can be implemented by the same arrangement as that of the delayed wave canceler  107 . 
   In order to perform signal estimation in the DDFSE  110 , an optimal timing must be determined. If only three components have certain amplitude values as shown in  FIG. 8 , it is not difficult to find a timing so as to set a 0  and a 1  in the MLSE region. If, however, eight components have certain amplitudes as shown in  FIG. 9 , the present invention requires the same function as that of the estimation region detector  203  in  FIG. 1 , which is used in the prior art. As this function, the estimation region detector  109  obtains an optimal timing based on the impulse response sequence newly obtained by the delayed wave canceler  107 . 
   The DDFSE  110  performs signal estimation upon receiving the impulse response sequence output from the delayed wave canceler  107 , the reception signal output from the delayed wave canceler  108 , and the timing signal output from the estimation region detector  109 . The estimation result is output as a decision result  111  from an output terminal T 3  (shown in  FIG. 5 ). 
   Only the preferred embodiment of the present invention has been exemplified above. However, the present invention is not limited to this. Persons skilled in the art easily recognize that various changes and modifications can be made within the spirit and scope of the invention.