Abstract:
Switch driver circuity having first and second output nodes with a current-voltage converter connected therebetween and providing current paths of first and second directions between the nodes, switching circuity connected therewith being switchable between first and second states respectively permitting current flow of a common preselected magnitude in respective first and second opposite directions producing potential differences between the first and second output nodes of a common magnitude but respective, opposite polarities.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to switch driver circuitry for use, for example, in digital-to-analog converters. 
     2. Description of the Related Art 
     FIG. 1 of the accompanying drawings shows parts of a previously-considered current-switched digital-to-analog converter (DAC)  1 . The DAC  1  is designed to convert an n-bit digital input word into a corresponding analog output signal. 
     The DAC  1  includes a plurality of individual binary-weighted current sources  2   1  to  2   n  corresponding respectively to the n bits of the digital input word applied to the DAC. Each current source passes a substantially constant current, the current values passed by the different current sources being binary-weighted such that the current source  2   1  corresponding to a least-significant-bit of the digital input word passes a current I, the current source  2   2  corresponding to the next-least-significant-bit of the digital input word passes a current  2 I, and so on for each successive current source of the converter. 
     The DAC  1  further includes a plurality of differential switching circuits  4   1  to  4   n  corresponding respectively to the n current sources  2   1  to  2   n . Each differential switching circuit  4  is connected to its corresponding current source  2  and switches the current produced by the current source either to a first terminal connected to a first connection line A of the converter or a second terminal connected to a second connection line B of the converter. The differential switching circuit receives one bit of the digital input word (for example the differential switching circuit  4   1  receives the least-significant-bit of the input word) and selects either its first terminal or its second terminal in accordance with the value of the bit concerned. A first output current I A  of the DAC is the sum of the respective currents delivered to the differential-switching-circuit first terminals, and a second output current I B  of the DAC is the sum of the respective currents delivered to the differential-switching-circuit second terminals. The analog output signal is the voltage difference V A -V B  between a voltage V A  produced by sinking the first output current I A  of the DAC  1  into a resistance R and a voltage V B  produced by sinking the second output current I B  of the converter into another resistance R. 
     FIG. 2 shows a previously-considered form of differential switching circuit suitable for use in a digital-to-analog-converter such as the FIG. 1 converter. 
     This differential switching circuit  4  comprises first and second PMOS field effect transistors (FETs) S 1  and S 2 . The respective sources of the transistors S 1  and S 2  are connected to a common node TAIL to which a corresponding current source ( 2   1  to  2   n  in FIG. 1) is connected. The respective drains of the transistors S 1  and S 2  are connected to respective first and second output nodes OUTA and OUTB of the circuit which correspond respectively to the first and second terminals of each of the FIG. 1 differential switching circuits. 
     Each transistor S 1  and S 2  has a corresponding driver circuit  6   1  or  6   2  connected to its gate. Complementary input signals IN and INB are applied respectively to the inputs of the driver circuits  6   1  and  6   2 . Each driver circuit buffers and inverts its received input signal IN or INB to produce a switching signal SW 1  or SW 2  for its associated transistor S 1  or S 2  such that, in the steady-state condition, one of the transistors S 1  and S 2  is on and the other is off. For example, as indicated in FIG. 2 itself, when the input signal IN has the high level (H) and the input signal INB has the low level (L), the switching signal SW 1  (gate drive voltage) for the transistor S 1  is at the low level L, causing that transistor to be ON, whereas the switching signal SW 2  (gate drive voltage) for the transistor S 2  is at the high level H, causing that transistor to be OFF. Thus, in this condition, all of the input current flowing into the common node TAIL is passed to the output node OUTA and no current passes to the output node OUTB. 
     When it is desired to change the state of the circuit  4  of FIG. 2 so that the transistor S 1  is OFF and the transistor S 2  is ON, complementary changes are made simultaneously in the input signals IN and INB such that the input signal IN changes from H to L at the same time as the input signal INB changes from L to H. As a result of these complementary changes, it is expected that the transistors S 1  and S 2  will switch symmetrically, that is that the transistor S 1  will turn OFF at exactly the same moment that the transistor S 2  turns ON. However, in practice there is inevitably some asymmetry in the turn-ON and turn-OFF speeds. This can result in a momentary glitch at the common node TAIL which may in turn cause glitches at one or both output nodes of the circuit, producing a momentary error in the DAC analog output value until all of the switches have switched completely. These glitches in the analog output signal may be code-dependent and result in harmonic distortion or even non-harmonic spurs in the output spectrum. 
     As the size of the glitch associated with the switching of the differential switching circuit is dependent on the symmetry of the complementary changes in the input signals IN and INB, much attention has been directed to generating and delivering these input signals to the differential switching circuit synchronously with one another. However, it is found in practice that, even if the input signals are perfectly symmetrical, the drive circuits  6   1  and  6   2  which derive the switching signals from the input signals inevitably introduce asymmetry into the switching signals SW 1  and SW 2  which actually control the transistors S 1  and S 2 . Such asymmetry results in transient output current distortion in any individual differential switch circuit. Furthermore, in a DAC employing multiple differential switch circuits, it also results in a variation between the switching times of the different circuits. These variations lower the spurious-free dynamic range (SFDR) of the DAC (a measure of the difference, in dB, between the rms amplitude of the output signal and the peak spurious signal over the specified bandwidth). These variations also lead to code-dependency of the analog output signal of the converter. 
     SUMMARY OF THE INVENTION 
     According to a first aspect of the present invention there is provided switch driver circuitry comprising: first and second output nodes; a current-voltage converter connected to said first and second output nodes to provide a current path through which current is permitted to flow in a first direction from said first to said second output node, or in a second direction from said second to said first output node, when the circuitry is in use, for producing a potential difference between said first and second output nodes that is dependent upon the magnitude and direction of the current flow; and switching circuitry connected with said first and second output nodes and switchable, in dependence upon an applied control signal, from a first state, in which a current of preselected magnitude is caused to flow in said first direction through said current path, to a second state in which a current of substantially the same magnitude as said preselected magnitude is caused to flow in said second direction through said current path, a current-voltage characteristic of the current-voltage converter being such that said potential differences produced respectively in said first and second states have substantially the same magnitudes but opposite polarities. 
     Such switch driver circuitry can provide improved symmetry of operation. 
     According to a second aspect of the present invention there is provided a switch circuit comprising: first and second output nodes; a current-voltage converter connected to said first and second output nodes to provide a current path through which current is permitted to flow in a first direction from said first to said second output node, or in a second direction from said second to said first output node, when the circuitry is in use, for producing a potential difference between said first and second output nodes that is dependent upon the magnitude and direction of the current flow; switching circuitry connected with said first and second output nodes and switchable, in dependence upon an applied control signal, from a first state, in which a current of preselected magnitude is caused to flow in said first direction through said current path, to a second state in which a current of substantially the same magnitude as said preselected magnitude is caused to flow in said second direction through said current path, a current-voltage characteristic of the current-voltage converter being such that said potential differences produced respectively in said first and second states have substantially the same magnitudes but opposite polarities; a first switch element having a control terminal connected to said first output node and switchable from an OFF state to an ON state by the change in the first-output-node potential brought about when said switching circuitry is switched from one of said first and second states to the other of those states; and a second switch element having a control terminal connected to said second output node and switchable from an ON state to a OFF state by the change in the second-output-node potential brought about when said switching circuitry is switched from said one state to said other state. 
     According to a third aspect of the present invention there is provided a digital-to-analog converter comprising switch driver circuitry comprising: first and second output nodes; a current-voltage converter connected to said first and second output nodes to provide a current path through which current is permitted to flow in a first direction from said first to said second output node, or in a second direction from said second to said first output node, when the circuitry is in use, for producing a potential difference between said first and second output nodes that is dependent upon the magnitude and direction of the current flow; switching circuitry connected with said first and second output nodes and switchable, in dependence upon an applied control signal, from a first state, in which a current of preselected magnitude is caused to flow in said first direction through said current path, to a second state in which a current of substantially the same magnitude as said preselected magnitude is caused to flow in said second direction through said current path, a current-voltage characteristic of the current-voltage converter being such that said potential differences produced respectively in said first and second states have substantially the same magnitudes but opposite polarities; the digital-to-analog converter further comprising: a first switch element having a control terminal connected to said first output node and switchable from an OFF state to an ON state by the change in the first-output-node potential brought about when said switching circuitry is switched from one of said first and second states to the other of those states; a second switch element having a control terminal connected to said second output node and switchable from an ON state to a OFF state by the change in the second-output-node potential brought about when said switching circuitry is switched from said one state to said other state, said first switch element being connected between first and second converter nodes and said second switch element being connected between said first node and a third converter node; and a current source or current sink connected operatively to said first converter node for causing a substantially constant current to pass through said first converter node when the converter is in use. 
     According to a fourth aspect of the present invention there is provided a digital-to-analog converter comprising: a plurality of differential switching circuits, each differential switching circuit having switch driver circuitry comprising: first and second output nodes; a current-voltage converter connected to said first and second output nodes to provide a current path through which current is permitted to flow in a first direction from said first to said second output node, or in a second direction from said second to said first output node, when the circuitry is in use, for producing a potential difference between said first and second output nodes that is dependent upon the magnitude and direction of the current flow; switching circuitry connected with said first and second output nodes and switchable, in dependence upon an applied control signal, from a first state, in which a current of preselected magnitude is caused to flow in said first direction through said current path, to a second state in which a current of substantially the same magnitude as said preselected magnitude is caused to flow in said second direction through said current path, a current-voltage characteristic of the current-voltage converter being such that said potential differences produced respectively in said first and second states have substantially the same magnitudes but opposite polarities; each said differential switching circuit further having: a first switch element having a control terminal connected to said first output node and switchable from an OFF state to an ON state by the change in the first-output-node potential brought about when said switching circuitry is switched from one of said first and second states to the other of those states; a second switch element having a control terminal connected to said second output node and switchable from an ON state to a OFF state by the change in the second-output-node potential brought about when said switching circuitry is switched from said one state to said other state, said first switch element being connected between first and second nodes of the differential switching circuit and said second switch element being connected between said first node and a third node of the differential switching circuit; and the respective second nodes of the differential switching circuits of said plurality being connected together, and the respective third nodes of the differential switching circuits of said plurality being connected together; and the digital-to-analog converter further comprising a plurality of current sources or current sinks, corresponding respectively to the differential switching circuits of said plurality, each current source or current sink being connected operatively to said first node of its said corresponding differential switching circuit for causing a substantially constant current to flow therethrough when the converter is in use. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1, discussed hereinbefore, shows parts of a previously-considered current-switched DAC; 
     FIG. 2 shows parts of previously-considered switch driver circuitry in the FIG. 1 DAC; 
     FIG. 3 shows parts of switch driver circuitry according to a first embodiment of the present invention; 
     FIG. 4 shows an example of current switching circuitry to which the FIG. 3 embodiment can be connected; 
     FIGS.  5 (A) to  5 (D) show operating waveforms generated by the FIG. 3 embodiment when in use; 
     FIGS.  6 (A) and  6 (B) are diagrams for use respectively in explaining operation of the FIG. 3 embodiments in first and second different states; 
     FIG. 7 shows a graph for use in explaining a current-voltage characteristic of a circuit element in the FIG. 3 embodiment; 
     FIG. 8 shows a modification which can be applied to embodiments of the invention; and 
     FIG. 9 shows parts of switch driver circuitry according to a second embodiment of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     FIG. 3 shows parts of switch driver circuitry according to a preferred embodiment of the present invention. The circuitry  10  includes respective first and second inverting input buffers  12  and  14 . The first input buffer receives at an input thereof a first input signal IN and the second input buffer  14  receives at an input thereof a second input signal INB complementary to the first input signal IN. The first input buffer  12  inverts the received IN signal to produce at an output thereof an inverted signal INVB. Similarly, the second input buffer  14  inverts the received INB signal to produce at an output thereof an inverted signal INV. The signals IN, INB, INV and INVB are all logic signals which change between a high logic level (H) and a low logic level (L). 
     The inverted signal INVB is supplied from the output of the first input buffer  12  to an input of a first inverting output buffer  16 . As shown in FIG. 3, the output buffer  16  includes respective PMOS FET and NMOS FET transistors  18  and  20 . The PMOS FET transistor  18  has its source connected to a first common node CN 1  of the circuitry, its gate connected to the output of the first input buffer  12  and its drain connected to a first output node ON 1  of the circuitry. The NMOS FET  20  has its source connected to the first output node ON 1 , its gate connected to the output of the first input buffer  12 , and its drain connected to a second common node CN 2  of the circuitry. 
     The circuitry also includes a second inverting output buffer  22  which, like the first output buffer  16 , has respective series-connected PMOS FET and NMOS FET transistors  24  and  26 . The PMOS FET  24  has its source connected to the first common node CN 1 , its gate connected to the output of the second input buffer  14 , and its drain connected to a second output node ON 2  of the circuitry. The NMOS FET  26  has its source connected to the second output node ON 2 , its gate connected to the output of the second input buffer  14 , and its drain connected to the second common node CN 2 . 
     Connected between a positive supply line ANALOG VDD and the first common node CN 1  of the circuitry are a constant current source transistor  28  and a cascode transistor  30 . Each of the transistors  28  and  30  is a PMOS FET. The constant current source transistor  28  has its gate connected to a first biassing line B 1  of the circuitry which, in use of the circuitry, is maintained at a potential V pcs  that is fixed relative to the potential of the positive supply line ANALOG VDD. The cascode transistor  30  has its gate connected to a second biassing line B 2  of the circuitry which, in use of the circuitry, is maintained a potential V pcasc  which is also fixed in relation to ANALOG VDD potential. 
     Connected between the second common node CN 2  of the circuitry and a ground potential supply line GND of the circuitry are series-connected first and second resistors R 1  and R 2  and, in parallel with the resistors, a capacitor C 1 . The resistors R 1  and R 2  have a total resistance of approximately 5 kΩ in this embodiment, with a 1:2 resistance ratio. The capacitor C 1  has a capacitance of, for example, 100 fF in this embodiment. 
     Connected between the first and second output nodes ON 1  and ON 2  of the circuitry  10  is a further PMOS FET  32 . The PMOS FET  32  has first and second current-path terminals connected respectively to the first and second output nodes ON 1  and ON 2 . One of the first and second current-path terminals is the source of the FET and the other of the current-path terminals is the drain of the FET, the source and drain designations being dependent on the in-use potentials of the output nodes. Following convention, the higher-potential current-path terminal for a PMOS FET is designated the source, and the lower-potential current-path terminal is designated the drain. As will be explained hereinafter, these designations are swapped around in use of the circuitry. The gate of the transistor  32  is connected to a junction node JN between the first and second resistors R 1  and R 2 . 
     As shown in FIG. 4, the FIG. 3 circuitry may be used to drive current switching circuitry of the same kind as described already with reference to FIG.  2 . Accordingly, a description of the current switching circuitry is not repeated here. The first main switching transistor S 1  in FIG. 4 has its gate connected to the first output node ON 1  of the FIG. 3 switch driver circuitry, and the second main switching transistor S 2  in FIG. 4 has its gate connected to the second output terminal ON 2  of the FIG. 3 switch driver circuitry. As indicated by the parts shown with dotted lines in FIG. 4, each branch of the current switching circuitry preferably includes a cascode transistor  42  or  44  connected between the main switching transistor S 1  or S 2  of the branch and the output terminal OUTA or OUTB of the branch. These optional cascode transistors are described more fully in our co-pending U.S. patent application Ser. No. 09/634,588 (corresponding to United Kingdom patent application no. 9926653.8), the entire content of which is incorporated herein by reference. The cascode transistor  42  or  44  in each branch has its source connected to the drain of the main switching transistor S 1  or S 2  of the branch concerned, its gate connected to the ground potential supply line GND, and its drain connected to the output terminal OUTA or OUTB of the branch concerned. 
     Operation of the FIG.  3  and FIG. 4 circuitry will now be described with reference to FIGS.  5 (A) to  5 (D) and  6 (A) and  6 (B). Incidentally, to make the timing relationships between the various signals easier to see in FIGS.  5 (A) to  5 (D), FIG.  5 (B) is repeated as FIG.  5 (C). 
     Initially, i.e. prior to time A in FIGS.  5 (A) to  5 (D), the first input signal IN has the low logic level L, and the second input signal INB has the high logic level H. This means that the inverted signals INVB and INB are H and L respectively. In this condition, as shown in FIG.  6 (A), in the first output buffer  16  the PMOS FET  18  is OFF and the NMOS FET  20  is ON. In the second output buffer  22 , the PMOS FET  24  is ON and the NMOS FET  26  is OFF. 
     The constant current source transistor  28  supplies a substantially constant current I from the positive supply line ANALOG VDD to the first common node CN 1 . The current I is, for example, 150 μA. The current I passes through the cascode transistor  30  which serves to shield the drain of the current source transistor  28  from voltage fluctuations caused by fluctuations in the potential of the first common node CN 1  arising in use of the circuitry. 
     Thus, the current I supplied to the first common node CN 1  has a first path P 1  between the first and second common nodes, as shown in FIG.  6 (A). This path passes (in order) through the channel of the PMOS FET  24 , the second output node ON 2 , the channel of the PMOS FET  32 , the first output node ON 1 , and the channel of the NMOS FET  20 . From the second common node CN 2 , the current I then passes through the resistor R 1 , the junction node JN and the second resistor R 2 , to reach the ground potential reference line GND. 
     The potentials generated at the various circuitry nodes in this condition are as follows (see FIG.  5 (B)). The potential V JN  of the junction node JN is determined by the product I.R 2  of the current I and the resistance of the second resistor R 2  which, in this embodiment, is approximately 0.36V. Similarly, the potential V CN2  of the second common node CN 2  is determined by I(R 1 +R 2 ) which, in this embodiment, is approximately 0.55V. The potential V ON1  of the first output node ON 1  is determined by the sum of the drain potential of the NMOS FET  20  and the on-state drain-source voltage of the NMOS FET  20 , i.e. V ON1 =V CN2 +V DS(ON)20 . In this embodiment, V DS(ON)20  is approximately 50 mV, so that V ON1  is approximately 0.60V. 
     The current I flows through the PMOS FET  32  from the second output node ON 2  to the first output node ON 1 . This means that the source of the transistor  32  (i.e. its higher-potential current-path terminal) is connected to the second output node ON 2 , and its drain is connected to the first output node ON 1 . The current I flowing through the transistor  32  is set high enough to place the transistor  32  in a saturated operating region. In this case, the gate-source voltage V GS32  of the transistor  32  has an unique value determined by the current density in the transistor  32 , i.e. V GS32 =V TP −(I/k), where I is the current flowing through the transistor  32  and V TP  and k are parameters of the transistor  32  determined by its physical structure. 
     For example, V GS32  is approximately −0.9V in this embodiment. To obtain the source potential of the transistor  32  it is necessary to subtract this gate-source voltage V GS32  from the gate voltage of the transistor  32 . This source potential of the transistor  32  determines the potential V ON2  of the second output node. Thus, V ON2 =V JN −V GS32 . In this embodiment, with V JN ≈0.36V and V GS32 ≈−0.90V, V ON2  is approximately equal to 1.25V. 
     The potential V CN1  of the first common node CN 1  is determined by the source potential of the PMOS FET  24 . This source potential is in turn determined by the drain potential of the PMOS FET  24 , i.e. V ON1 , and the ON-state drain-source voltage V DS(ON)24  of the PMOS FET  24 . Thus, V CN1 =V ON2 −V DS(ON)24 . Typically, V DS(ON)24  is approximately −150 mV, so that V CN1  is approximately equal to 1.40V in this embodiment. 
     In this condition (FIG.  6 (A)) the first output node ON 1  has a predetermined ON output potential V on  of the circuitry, and the second output node ON 2  has a predetermined OFF output potential V OFF  of the circuitry, i.e. V ON1 =V on  and V ON2 =V off . In this embodiment, V on ≈0.60V and V off ≈1.25V. When these potentials are applied to the switching transistors S 1  and S 2  in the current switching circuitry, the transistor S 1 , which receives the ON output potential V on , is turned ON, and the switching transistor S 2 , which receives the OFF output potential V off , is turned OFF. As a result, the potential difference V B −V A  between the output terminals OUTB and OUTA is negative, as shown in FIG.  5 (D). 
     Incidentally, the other potential differences V CASCB −V CASCA  and V B ′−V A ′ shown in FIG.  5 (D) are internal signals within the current switching circuitry and will not be discussed further here. 
     At time A in FIGS.  5 (A) to  5 (D) the first and second input signals IN and INB undergo respective complementary logic level changes (L to H for IN, and H to L for INB). In response to these changes the input buffer output signals INV and INVB also undergo complementary logic level changes (L to H for INV and H to L for INVB). As a result, as shown in FIG.  6 (B), a second current path P 2  between the common nodes CN 1  and CN 2  is created, different from the first current path P 1  shown in FIG.  6 (A). In this case, the current I supplied to the first common node CN 1  by the constant current source transistor  28  flows through the channel of the PMOS FET  18  in the first output buffer  16 , the first output node ON 1 , the PMOS FET  32 , the second output node ON 2  the and channel of the NMOS FET  26  in the second output buffer  22 . As in FIG.  6 (A), from the second common node CN 2  the current flows through the resistor R 1 , the junction node JN and the second resistor R 2 , before reaching the ground potential supply line GND. 
     After switching has taken place, it will be appreciated that the potentials V CN1  and V CN2  of the common nodes are substantially unchanged from those prevailing before the switching took place, i.e. the potentials of the common nodes are the same in FIGS.  6 (A) and  6 (B). This is because the same current I flows through the second current path P 2  in FIG.  6 (B) as flows through the first current path P 1  in FIG.  6 (A). 
     Also, substantially the same ON and OFF output potentials V on  and V off  are generated in FIG.  6 (B) as were generated in FIG.  6 (A). In FIG.  6 (B), however, the ON output potential V on  is generated at the second output node ON 2 , and the OFF output potential is generated at the first output node ON 1 , i.e. V ON1 =V off  and V ON2 =V on . 
     It will also be appreciated that in FIG.  6 (B), the same current I flows through the transistor  32  as flowed in the FIG.  6 (A) case, but in the opposite direction, i.e. from the first output node ON 1  to the second output node ON 2  in FIG.  6 (B). The current-voltage characteristic of the transistor  32  is shown in FIG.  7 . In FIG. 7, the vertical axis represents current flowing through the transistor channel, and the horizontal axis represents the potential difference between the first and second current-path terminals (i.e. the potential difference across the transistor channel). As can be seen from FIG. 7, the I-V characteristic is perfectly symmetrical for both positive and negative values of the current flowing through the transistor, i.e. whichever direction the current is flowing. This means that the potential difference ΔV between the ON and OFF output potentials in FIGS.  6 (A) and  6 (B) is exactly the same. Furthermore, during switching, the potentials at the first and second output nodes ON 1  and ON 2  of the circuitry have the same rising and falling waveforms when switching (at time A) from the state shown in FIG.  6 (A) to the state shown in FIG.  6 (B) as when switching (at time B) from the state shown in FIG.  6 (B) to the state shown in FIG.  6 (A). This effect can clearly be seen from a comparison of the waveforms at times A and B in FIG.  5 (B). 
     The FETs  18 ,  20 ,  24  and  26  in the output buffers  16  and  22  are desirably very small to provide for fast switching. As a consequence of their small sizes, they tend not to be closely matched. The implications of the mismatches in terms of both delay variation and amplitude variation of the ON and OFF potentials will now be considered. 
     In terms of delay variation, because the FETs in the switch driver circuitry are very small the rise and fall times of the output node potentials are very fast (see FIG.  5 (B)). This means that although there will be delay mismatches between the FETs of the switch driver circuitry, the magnitude of the resulting delay variation at the output nodes is also very small. 
     In terms of amplitude variation the PMOS FETs  18  and  24  do not influence the output potentials, and so if they are not matched there is no significant impact on the symmetry of the output potentials. The NMOS FETs  20  and  26  affect the output potentials only weakly (because although V on  is influenced by V DS(ON)  of the NMOS FET  20  or  26  that is on, V DS(ON)  is itself small, e.g. 50 mV). The ON and OFF output potentials therefore only have a very small asymmetry due to mismatches of the transistors in the output buffers. 
     The capacitor C 1  is a decoupling capacitor provided to make the potential V TAIL  in the current switching circuitry settle as fast as possible. Referring to FIG.  5 (B) it can be seen that when switching occurs, V TAIL  has a small rise. This rise is caused by the transient at the second common node CN 2  that occurs during switching. In order to make V TAIL  settle as quickly as possible it is desirable to reduce the CN 2  transient. This is achieved, at the expense of a larger transient at the first common node CN 1 , by means of the capacitor C 1  coupled between CN 1  and GND. The transient on CN 1  does not affect the current switching circuitry, and is therefore insignificant. The capacitance value is preferably set to provide a time constant of around 500 ps, similar to the settling times of the internal signals of the switch driver circuitry. Thus, when the sum of R 1  and R 2  is approximately 5 kΩ, C 1  should have a capacitance of approximately 100 fF (giving a RC time constant of 500 ps). 
     The transistor  32  also provides the following further advantages. Firstly, as it has a non-linear I-V characteristic, the voltage developed across it is relatively large even when the current flowing through the channel is relatively low, as occurs during switching (i.e. before and after the crossover of the rising and falling waveforms in FIG.  5 (B). This leads to a very fast settling time for the output node potentials after switching, because most of the switch driver current I is available for driving the output nodes rather than being wasted in the transistor  32 . For example, in FIG.  5 (B) it can be seen that the rising waveform, which is slower than the falling waveform, settles in approximately 600 ps. Thus, in the FIG. 3 switch driver circuitry, all of the internal signals settle in less than 600 ps. The effect of applying these fast-settling internal signals to the FIG. 4 current switching circuitry is illustrated in FIG.  5 (D). In FIG.  5 (D), it is assumed that the cascode transistors  42  and  44  are present. The resulting rise time of the potential difference between the output terminals OUTA and OUTB is approximately 350 ps (for the rise from 10% to 90% of full-scale value). This can provide an output bandwidth of 1 GHz, facilitating a typical sampling rate F DAC  of the DAC of 1.6 G samples/s with a worst-case rate of 1 G samples/s. 
     The second advantage is that, because the transistor  32  is a PMOS FET like the transistors in the current switching circuitry of FIG. 4, its saturation drain-source voltage V DS(SAT)  varies in the same way as the drain-source saturation voltages V DS(SAT)  of the transistors in the current switching circuitry. This is important, as in practice, the drain-source saturation voltage V DS(SAT)  of a PMOS transistor may vary by a factor of 2 due to process and/or temperature variations. 
     Considering the FIG. 4 current switching circuitry in more detail, at any given time, one of the main switching transistors S 1  and S 2  is OFF and the other is ON. Referring to FIG.  6 (B), for the purposes of explanation it will be assumed that the OFF transistor is the transistor S 1  and the ON transistor is the transistor S 2 . In this condition, the potential V TAIL  of the sources of the transistors S 1  and S 2  is influenced by the drain-source potential of the ON transistor S 2 . When the switching transistors S 1  and S 2  have a relatively high drain-source saturation voltage V DS(SAT)S VTAIL  is increased as compared to when V DS(SAT)S  is low. This means that in order to maintain the OFF transistor S 1  in the OFF condition, its gate voltage, i.e. the OFF potential V OFF , must also be increased. This increase occurs automatically in the FIG. 3 switch driver circuitry because in that circuitry the difference between the OFF and ON potentials is increased when the drain-source saturation voltage V DS(SAT)32  of the transistor  32  is relatively high as compared to when that drain-source saturation voltage is relatively low. Accordingly, the OFF potential is self-regulating in the FIG. 3 switch driver circuitry. 
     In the FIG.3 circuitry it is also desirable to make the ON output potential track V DS(SAT)32  of the switching transistors S 1  and S 2  and the cascode transistors  42  and  44  (if used) in the current switching circuitry. Considering FIG.  6 (A), and assuming the cascode transistors are present, in the branch of the current switching circuitry that is on, the ON output potential V on  must be sufficient for both the cascode transistor  42  and the switching transistor S 1  to be maintained in the saturated condition, even when V DS(SAT)  of each of those transistors varies. The nominal drain-source saturation voltage V DS(SAT)S  of the switching transistors is, for example, 200 mV. The nominal drain-source saturation voltage V DS(SAT)C  of the cascode transistors is, for example 300 mV. By setting V on  to a nominal value of 0.6V the potential difference between the cascode transistor gate (GND) and the switching transistor gate (V on ) exceeds V DS(SAT)C  by 1.5 times the nominal V DS(SAT)S  of the switching transistor. However, as V DS(SAT)S  and V DS(SAT)C  can each vary by a factor of 2 with process/temperature, preferably V on  should also increase when V DS(SAT)S  and/or V DS(SAT)C  increase. 
     This change in V on  to compensate for variations in V DS(SAT)S  of the switching transistors S 1  and S 2  (and for variations in V DS(SAT)C  of the cascode transistors  42  and  44 , if provided) can be achieved by making the resistances of the resistors R 1  and R 2  in the FIG. 3 circuitry variable in dependence upon V DS(SAT)S  and/or V DS(SAT)C . One example of control circuitry for varying the resistances will now be described with reference to FIG.  8 . 
     In FIG. 8 the control circuitry  60  includes a first constant current source  62  connected between a positive power supply line ANALOG VDD of the circuitry and a first node N 1 . A first PMOS FET  64  has its source connected to the node N 1  and its gate and drain connected to the ground potential supply line GND. 
     The circuitry also includes a second PMOS FET  66  which has its source connected to the node N 1 . The gate and drain of the PMOS FET  66  are connected to a second node N 2 , and a constant current sink  68  is connected between the node N 2  and GND. 
     The current I 1  sourced by the constant current source  62  is large compared to the current I 2  sunk by the constant current sink  68 . Also, the first PMOS FET  64  is narrow compared to the second PMOS FET  66 . For example, the width of the FET  64  is w and the width of the FET  66  is 3 w, and I 1 =4I sw  and I 2 =I sw , where I sw  is the current which flows through each switching transistor S 1  or S 2  when ON. 
     The circuitry  60  further includes a high-output-resistance transconductance amplifier  70  having a first (negative) input connected to the node N 2 . A second (positive) input of the amplifier  70  is connected to a node N 3  of the circuitry. A second constant current source  72  is connected between the ANALOG VDD and the node N 3 . First and second NMOS FETs  74  and  76  are connected in series between the node N 3  and GND. The first NMOS FET  74  has its drain connected to the node N 3 , its gate connected to the output of the amplifier  70  and its source connected to the drain of the second NMOS FET  76 . The NMOS FET  76  has its gate connected to the output of the amplifier  70  and its source connected to GND. An output node N 4  of the circuitry  60  is connected to the output of the amplifier  70 . 
     To enable the resistances of the resistors R 1  and R 2  in the switch driver circuitry to be varied, the resistors R 1  and R 2  are implemented using respective first and second series-connected NMOS FET transistors  80  and  82 . The first NMOS FET  80  has its drain connected to the second common node CN 2  of the switch driver circuitry  10 , its gate connected to the output node N 4  of the control circuitry and its source connected to the junction node JN (gate of the transistor  32 ) in the switch driver circuitry  10 . The NMOS FET  82  has its drain connected to the junction node JN, its gate connected to the output node N 4  and its source connected to GND. In this embodiment the NMOS FET  80  has the same size as the NMOS FET  74  and the NMOS FET  82  has the same size as the NMOS FET  76 . Alternatively, there may be a predetermined scaling factor between the two FETS  74 / 80  and  76 / 82  of each pair. 
     The output node N 4  can also be connected to resistance-setting NMOS FETs in further segments of the DAC circuitry, so as to enable the control circuitry  60  to operate in common for all segments. 
     Operation of the FIG. 8 control circuitry will now be described. The elements  62  to  68  serve to generate at the node N 2  a potential V DS(SAT)P  which is a measure of the drain-source saturation voltage of the switching transistors in the current switching circuitry (FIG.  3 ). Because of the difference in currents flowing through the FETs  64  and  66 , and their different widths, the ratio of the current densities in the FETs  64  and  66  is 9:1 (=(I 1 -I 2 )/w:I 2 /3 w). Because V DS(SAT)  is proportional to the square root of current density, the ratio between the respective V DS(SAT) s of the FETs  64  and  66  is 3:1. The respective V T s of the FETs  64  and  66  are substantially the same. The potential at node N 1  becomes equal to V DS(SAT)64 +V T64 , where the drain-source saturation voltage V DS(SAT)64  of the FET  64  is e.g. 0.9V and the threshold voltage V T64  of the FET  64  is e.g. 1V. Thus, the potential V N1  of node N 1  is, for example, 1.9V. The voltage drop across the FET  66  is V DS(SAT)66 +V T66 , where V DS(SAT)66  is e.g. 0.3V and V T66  is e.g. 1V, i.e. 1.3V. Thus, the potential at node N 2  is approximately equal to V DS(SAT)64 −V DS(SAT)66 , and this potential is taken as the measure V DS(SAT)P  of drain-source saturation voltages of the switching and cascode transistors in the current switching circuitry. 
     Incidentally, because the measure V DS(SAT)P  is derived from the difference V DS(SAT)64 −V DS(SAT)66  between the respective V DS(SAT) s of two FETs  64  and  66 , it is possible that it will not accurately reflect the actual V DS(SAT) s of the FETs of interest in the current switching circuitry, i.e. the switching transistors and the cascode transistors (if used). However, if it is expected that the actual V DS(SAT) s of the FETs of interest will be, say, 0.6V in total, then it is preferable to set the conditions of the FETs  64  and  66  so that their respective V DS(SAT) s are offset equally on either side of that total actual V DS(SAT) , which is why in this example V DS(SAT)64  is set to 0.9V and V DS(SAT)66  is set to 0.3V. 
     The second constant current source  72  sources a current I 3  that in this embodiment is substantially equal to the current I sourced by the constant current source  24  in the switch driver circuitry of FIG.  3 . In this embodiment the NMOS FET  74  has the same (variable) resistance as the NMOS FET  80  is to provide the first resistor R 1 . Similarly, the second NMOS FET  76  has the same (variable) resistance as the NMOS FET  82  used to provide the resistor R 2 . This means that the voltage at the node N 3  is the same as the voltage V CN2  at the second common node CN 2  in the switch driver circuitry. The effect of the amplifier  70 , therefore, is to adjust the potential at the output node N 4  until the potential at the node N 3  is equal to the potential V DS(SAT)P  of the node N 2 . Changing the N 4 -node potential changes the potential at the node N 3  because the N 4 -node potential determines the respective resistances of the first and second NMOS FET transistors  74  and  76  in the control circuitry. 
     In this way, in this embodiment the potential V CN2  of the second common node CN 2  is set substantially equal to the measure V DS(SAT)P . 
     It will be appreciated that, in the FIG. 8 circuitry, the resistances of the resistors R 1  and R 2  (provided by the NMOS FETs  80  and  82 ) each vary in accordance with the potential at the node N 4 . Accordingly, as V CN2  is varied the potential variation at the junction node JN tracks the potential variation of the second common node CN 2  so as to maintain the gate potential of the transistor  32  as a substantially fixed proportion (e.g. ⅔) of the potential V CN2 . 
     The advantage of using the FIG. 8 control circuitry to adjust the potential of the second common node CN 2  is that the ON output potential V on  tracks V DS(SAT)  variations of the main switching transistors and (if used) the cascode transistors in the current switching circuitry. The PMOS FET  32  serves automatically to cause V OFF  to track V DS(SAT) . 
     It will also be appreciated that in place of the PMOS FET  32  in the FIG. 3 embodiment, other circuit elements can be connected between the first and second output nodes ON 1  and ON 2  of the circuitry to achieve the same basic current-voltage conversion effect. In each case, it is preferable that the circuit element used has the same I-V characteristic irrespective of the direction of current flow through the element concerned. The I-V characteristic of the circuit element is preferably non-linear so as to provide a higher resistance at low values of current and a lower resistance at high values of current, but a linear circuit element such as an ohmic resistance element could be used. 
     A second embodiment of the present invention, using an ohmic resistance element between the first and second output nodes, will now be described with reference to FIG.  9 . In FIG. 9, components that are the same as, or correspond closely to, components in the first embodiment of FIG. 3 have been denoted by the same reference numerals and an explanation thereof is omitted. 
     In the FIG. 9 embodiment, in place of the transistor  32 , a resistor  102  is connected between the first and second output nodes ON 1  and ON 2 . A further resistor  104  is connected between ANALOG VDD and the source of the constant current source transistor  28 . Also, a further resistor  106  is connected between the second common node CN 2  and GND in place of the series-connected resistors R 1  and R 2  in the first embodiment. Each of the resistors  102 ,  104  and  106  is an ohmic resistance element, for example a high-resistance n-diffusion resistor. 
     As in the first embodiment, the same current I that is sourced by the constant current source transistor  28  flows selectively either along a first current path P 1 , or along a second current path P 2 , through the circuitry, in dependence upon the state of the complementary input signals IN and INB. 
     As in the first embodiment, the potential V CN2  of the second common node is determined by the product of the current I and the resistance R 106  of the resistor  106 . In the second embodiment, the potential difference ΔV between the potentials of the first and second output nodes V ON1  and V ON2  is determined by the product of the current I and the resistance R 102  of the resistor  102 . The I-V characteristic of the resistor  102  is the same for both directions of current flow through it, so the potential difference ΔV is the same whichever state the circuitry is in (in the steady-state) 
     The resistor  104  is provided to cause the potential V S28  of the source of the current source transistor  28  to track changes in the resistance of the resistor  102 . Within the circuitry, the resistors  102  and  104  are preferably placed physically close to one another so that their resistances will have a substantially fixed ratio irrespective of variations in their resistances brought about by process and/or temperature variations. Such variations may exhibit “gradients” across the device in one or more directions as the segments are laid out in a certain pattern over the device substrate. The make the layout within each segment insensitive to such gradients (at least in one direction) the resistor  104  may be divided into 2 equally-sized portions on opposite sides respectively of the resistor  102 . This means that the resistor  104  has a common centroid with the resistor  102 . Then, if the resistance of the resistor  102  in a segment has an increased value, so will the resistance of the resistor  104  of that segment. This has the effect of lowering the potential V S28  at the source of the constant current source transistor  28  so that, assuming its gate potential V pcs  remains unchanged (relative to ANALOG VDD), its gate-source voltage is made less negative, thereby reducing the current I. In this way, the product I.R 102 , which defines ΔV, is left substantially unchanged despite the increase in R 102 . 
     The ratios of the resistances R 102 , R 104  and R 106  are, for example, 1:2:1, with I being approximately 80 μA and R 102  being approximately 7.5 kΩ. This provides a potential difference ΔV between the ON and OFF output potentials of approximately 0.6V. 
     When a resistance element such as the element  102  is used as the current-voltage conversion element it is not essential to use the matching resistance element  104  or, indeed, to carry out any compensation for resistance variation. In this respect, although the potential difference ΔV generated across the resistor  102  is kept substantially fixed by using such compensation, inevitably the change in current affects the circuitry in other ways and, for example, changes the speed of the switching operation of the segment. This may make it preferable to leave the current unchanged in response to resistance variations. 
     Comparing FIG. 4 with FIG. 9, a further advantage of the FIG. 4 circuitry over the FIG. 9 circuitry is that the resistance element  102  (and the compensating resistor  104  if used) is large physically compared to the PMOS FET  32 , because a suitably large resistance (e.g. 7.5 kΩ) can only be achieved with a large physical structure (HN resistors may have a resistance of 1 kΩ/square). Such large structures have an appreciable parasitic capacitance. Also, when resistances are used, scaling of the circuitry becomes difficult since, if (say) the current is halved, the resistances must be doubled to achieve the same voltage, whereas with the PMOS FET  32  the voltage across it is maintained when the transistor is halved in size. Even worse, when the resistance is doubled, parasitic capacitance is also doubled, so that compared to the half-size transistors the parasitic capacitance goes up by a factor of 4. This makes the PMOS FET  32  far more preferable to use as the current-voltage conversion element. 
     Although the use of a circuit element having the same I-V characteristic for both directions of current flow between the output nodes is preferable, it will be appreciated that, by using two closely-matched uni-directional circuit elements connected in parallel between the two output nodes, substantially the same effect can be achieved. For example, back-to-back diode elements could be employed between the two output nodes. Each diode could be implemented using an MOS transistor with its gate connected to its source. 
     Although the foregoing embodiments have employed p-channel switching transistors, it will be appreciated that the present invention can be applied in other embodiments to current switching circuitry employing n-channel switching transistors (and a current sink in place of the current source). In this case, the polarities of the supply lines and the conductivity types of the transistors in the switch driver circuitry are reversed. 
     Furthermore, although the present invention has been described in relation to DACs, it will be understood by those skilled in the art that the present invention is applicable to any type of circuitry that includes switch elements that need to switch in complementary manner with accurately-controlled complementary switching signals.