Abstract:
An impedance matching device that is particularly suited for high speed, high power applications is disclosed. The impedance matching devices employ magnetic coupling allowing their usage in a &#34;frequency-agile&#34; environments such as communication as well as communication jamming and deception techniques without creating high current, high voltage and thermal conditions that might otherwise damage the related switching devices. The impedance matching device provides a predetermined impedance to transform a load impedance Z L  into a desired characteristic impedance Z 0  over a frequency range having a lower and an upper band. The device comprises one or more first and second inductive components. Each of the first inductive components has an impedance characteristic associated with said lower band, whereas each of the second inductive component has an impedance characteristic associated with the upper band. The device further comprises switch means having an active and an inactive state and being of a predetermined number at least equal to the predetermined number of second components. Each of the switch means in its active state bypassing its corresponding second component and effectively shorting out its respective first component.

Description:
This is a continuation of parent application, Ser. No. 07/680,220 filed on Apr. 4, 1991, now abandoned. 
    
    
     FIELD OF THE INVENTION 
     The present invention is directed to a device for high power and high speed impedance matching. The invention is directly applicably to antenna couplers, tunable filters and other variable impedance matching requirements, and broadly applicable to all impedance matching requirements. The present invention is particularly suited for electronics warfare (EW) systems used for jamming and deception of &#34;frequency agile&#34; communication and guidance links, as well as &#34;frequency agile&#34; communication systems. 
     BACKGROUND OF THE INVENTION 
     Traditional impedance matching devices are usually based on either the &#34;π&#34; network or the &#34;T&#34; network, both of which are well known electronic circuits, and which are illustrated in FIGS. 1(a) and 1(b), respectively. 
     In the π network of FIG. 1(a) (so called because the network diagram resembles the Greek letter π), impedance matching is achieved by a single series impedance Z2 and two parallel impedances Z1 and Z3 to ground, one located at the input and one located at the output of Z2. Typically, series impedance Z2 is modelled as an inductor L, while parallel impedances Z1 and Z2 are modelled as capacitors C 1  and C 2 , respectively. The impedances may be fixed, but it is usually preferred that they be variable in order to give the circuit a range in frequencies over which the impedances may be matched. Thus, the π network of FIG. 1(a) is illustrated as comprising a series impedance in the form of a variable inductor and parallel impedances in the form of variable capacitors. 
     In the T network (so called because it resembles the letter T), impedance matching is achieved by using two series impedances Z1 and Z3 and a single parallel impedance Z2 to ground located at the node between Z1 and Z3. In the T network, the series impedances are also modelled as variable inductors L, and L 2  and the parallel impedance as a variable capacitor, as illustrated in FIG. 1(b). 
     The π and T networks form the building blocks for most conventional impedance matching circuits. They are well understood, can be modelled using existing computer-aided design methods, and can be used to form other, more complex impedance matching circuits. Although used in many conventional antenna coupler configurations, the π and T matching sections present certain drawbacks for applications, such as an antenna coupler. 
     If large magnitude impedance transformations are required, it is possible to develop extremely high RF potentials on one or more of the matching networks. For this reason it is often necessary to, in the case of a π section device, choose a capacitor or switch, for example, capable of withstanding voltages in excess of 10,000 volts. In some cases, this voltage can ionize the air and cause a shorting path. For this reason, vacuum capacitors and relays such as those manufactured by the Jennings Corporation are often used. Solid state circuits matching circuits based on a π or T configuration can expose the solid state switches to extreme current and voltage conditions especially when attempting to match a &#34;short antenna&#34; at the low end of the HF frequency range (HF is typically 2 to 30 MHz). For example, when using a π section for matching to a 15 foot monopole antenna, a current as high as 10 amperes may flow through the inductor Z 2 , when attempting to transmit 1 to 2 kw power due to the typical impedance of this antenna varying from 0.5-j 900 ohms at 2 MHz to 400+j 500 ohms at 15 MHz. The inductor of the matching network may be shorted-out during a portion of this frequency range in order to obtain the needed impedance matching which, in turn, may cause its related solid state switch to experience a current as high as 20 amperes for this short duration. Furthermore, the voltage across the inductor at the 2 megahertz frequency may be in excess of 10 KV, and unless each turn forming the inductor is switched separately, the required voltage rating for the solid state switches, in their off-state, may be substantially over 1 KV. These high current and voltage conditions caused by the π as well as the T matching sections impose high stress on the solid state devices. Also, because it is desired to mount the switches in close proximity to the inductor for high frequency applications in order to overcome the disadvantages of switch lead inductance, a serious thermal problem may arise because of the heat transferred from the inductor to the solid-state switch. These high voltage, current, and thermal conditions may contribute to the premature failure of the solid state devices. 
     There is a need for an impedance matching device that overcomes the drawbacks of the conventional impedance matching π or T circuits. In particular, there is a need for an impedance matching device that reduces or even eliminates the high voltage stresses that are placed on the switches used at high power and high frequency applications to prevent the use of semiconductor switches such as diodes or the switch described in U.S. Pat. No. 4,808,859. Further, the impedance matching device should perform over a wide range of frequencies so that the related antenna or other device can be properly matched to the desired impedance. 
     In addition to overcoming the drawbacks of the π and T matching sections, there is a need to minimize the number of solid state switches associated with impedance matching devices. The numbers of switch devices are directly related to the number of switchable segments of the matching sections, wherein one switching device operatively connects or disconnects a corresponding switchable segment. It is desired that means be provided to reduce the number of switching devices needed for impedance matching devices. 
     Accordingly, it is an object of the present invention to provide impedance matching devices that reduce the number of switching devices, especially for high power and high frequency applications. 
     It is another object of the present invention to provide impedance matching devices that overcome the drawbacks of conventional impedance matching π or T circuits related to the high current, voltage and thermal stress conditions of the switching devices. 
     SUMMARY OF THE INVENTION 
     The present invention is directed to a device that is particularly suited for providing high power and high speed impedance matching and which translate the stress conditions on the related switching devices to values that can be managed. The impedance matching device provides a variable impedance over a frequency range having a lower and an upper band, and which transforms a high, unpredictable, or changing load impedance Z L  into a desired source impedance Z 0  for maximum power transfer. The impedance matching device comprises one or more first planar coupled inductive components, one or more second planar inductive components, and one or more switch means each with an active and an inactive state. Each of the one or more switch means is operatively connected to a corresponding second component and the number of such switch means equals at least the number of second components. The one or more second components can each have a predetermined impedance characteristic related to the frequency range. Each of the second components faces and is spaced apart from a respective second component. The first and second respective components are magnetically and electrically coupled to each other. Each of the switch means in its active state causes its corresponding second component to be &#34;bypassed&#34; and also effectively shorts out the respective first component. 
    
    
     DESCRIPTION OF THE DRAWINGS 
     For the purpose of illustrating the invention, there is shown in the drawings a form which is presently preferred; it being understood, however, that this invention is not limited to the precise arrangements and instrumentalities shown. 
     FIGS. 1(a) and 1(b) respectively illustrate the prior art π and T impedance matching circuits already described with reference to the &#34;Background&#34;. 
     FIG. 2 illustrates the basic principle of a prior art variable-length transmission line (VTL) used for impedance matching applications. 
     FIG. 3 illustrates the basic principle of a variable-length transmission line (VTL) that may be formed with the impedance matching device of the present invention. 
     FIG. 4 illustrates a second planar inductive component of an impedance matching circuit according to the invention. 
     FIG. 5 illustrates a first planar inductive component of an impedance matching circuit according to the invention. 
     FIG. 6 illustrates another embodiment of the second planar inductive component having multiple windings. 
     FIG. 7 is a model representation of one of the impedance matching circuits of the present invention. 
     FIG. 8 illustrates an impedance matching device that provides for an extended frequency range suitable for compact rearrangement. 
     FIG. 9 illustrates the impedance matching device of FIG. 8 arranged in a more compact manner denoted in this invention as a p-section. 
     FIG. 10 illustrates cascaded p-section as the impedance matching device of the present invention arranged as a multi-layer stripline arrangement, denoted also as a multi-layer magnetically coupled suspended stripline (MMCSS). 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     The present invention relates to an improved impedance matching device as well as various circuit arrangements that are particularly suited for high power and high frequency applications. The present invention provides for impedance matching and utilizes a variable length transmission line (VTL) technique. The present invention may be more fully appreciated by first describing the basic principle of the variable length transmission line (VTL) illustrated in FIG. 2. 
     FIG. 2 illustrates the imposition of an impedance matching arrangement 10 between a source of excitation 12 and a load impedance Z L . The source 12 may be of the type discussed with regard to the &#34;Background&#34; and is of a known frequency, typically but not limited to the frequency range from 2 to 30 megahertz (MHz), but it can be scaled in frequency to any desired frequency range. The source 12 is of a relatively high power, typically of several kilowatts (kW). The impedance Z L  may represent that of a &#34;short&#34; antenna which is operated over the 2-30 MHz frequency range but it can be &#34;any&#34; arbitrary antenna or other device impedance. The circuit arrangement 10 transforms the impedance Z L  into a desired characteristic impedance Z O  so that the power of the source 12 may be substantially absorbed by the load impedance. The circuit 10 comprises a plurality of impedance line segments 14 1 , 14 2 , 14 3 , 14 4 , 14 N-3 , 14 N-2 , 14 N-1  and 14 N , each having a respective characteristic impedance Z 01 , Z 02 , . . . Z 0N , and each respectively operatively connected by a switch SW 1 , SW 2 , SW 3 , SW 4 , SW N-3 , SW N-3 , SW N-1  and SW N  arranged as shown in FIG. 2. The symbol illustrating a broken line for segments 14 1  . . . 14 N  is used in FIG. 2 to indicate that the line length of these segments is a selectable design parameter. This broken line symbol is used in other Figures to indicate the same function. The switches SW 1 , SW 2 , . . . SW N  of FIG. 2 are operatively responsive to a switch control module 16 which selectably closes one or more switches SW 1 , . . . SW N  according to a preselected program which provides for desired impedance matching values that accommodate various load impedances Z L . 
     FIG. 2 illustrates the basic principle of the variable-length transmission line (VTL) whose overall line length is controlled by the operation of the switches SW 1 , . . . SW N  connecting or disconnecting respective segments 14 1 , . . . 14 N . Each of the switches SW 1 , . . . SW N  has an active state which shorts-out the transmission sections 14 1 , 14 N , respectively, and in its inactive state allows the respective transmission section 14 1 , . . . 14 N  to be selectively interposed between Z L  and the source 12. The length of the transmission line interposed, that is connected between source 12 and Z L  may vary from a &#34;zero length&#34;, commonly referred to as l min , to a maximum length, commonly referred to as l max  The zero length is provided when all of the switches SW 1 , . . . SW N  are in their active states so as to short out all of the respective transmission line 14 1 , . . . 14 N , whereas, the l max   is realized when all of the switches SW 1 , . . . SW N  are in their inactive state so as to serially interpose all of the line segments 14 1 , . . . 14 N  between the source 12 and the load Z L . The incremental changes, that may be realized in the length of the variable transmission line 14 1 , . . . 14 N  are directly dependent upon the number of transmission lines 14 1 , . . . 14 N  and the number of switches SW 1 , . . . SW N  that are employed. The present invention reduces the number of switches (SW K  . . . SW N ) by a factor of at least about one-half while still providing the desired impedance matching achievable by the arrangement of FIG. 2. 
     FIG. 3 illustrates an impedance matching device 20 according to the present invention for providing a predetermined impedance over a frequency range having a lower and an upper band. The impedance device 20 comprises one or more first planar inductive components shown as 22a, 22b, . . . 22 K-1 , and 22 K . FIG. 3 shows the components 22a, 22 b1 , 22 K-1 , and ... 22 K  as comprised of loops. Each loop may have as many as 7 turns, where γ is an integer of at least one (1) or greater. The first planar inductive components 22 a  . . . 22 K  are herein termed &#34;primary loops or windings&#34;. Each loop of the primary winding 22 a  . . . 22 K , consisting of γ turns, faces one overlapping large loop (i.e., one turn) of a secondary loop or winding respectively shown in FIG. 3 as 24 a , 24 b , 24 K-1 , and 24 K . The winding 24 a  . . . 24 K  are also termed herein the &#34;second planar inductive components&#34;. The subscript K may be defined as: 
     
         K=N/γ                                                (1) 
    
     where N is the number of transmission segment 14 shown in FIG. 2, and 7 is the number of overlapping primary loops (22 a  . . . 22 K ) facing a respective secondary loop (24a . . . 24 K ). The first components 22a, . . . 22 K  are respectively associated with second components 24a, . . . 24 K . Each of the first component 22 a , . . . 22 K  and each of the second components 24 a , . . . 24 K  have a characteristic impedance selected for matching purposes. The device 20 further comprises a plurality of switch means (SW), shown in FIG. 3 as SW 0 , SW 1 , SW 2 , SW K-1 , SW K  and SW K+1 , each having an active and inactive state and being of a number in which may be expressed as: 
     
         μ=K+2                                                   (2) 
    
     The first planar components 22a, 22b . . . 22 K  are connected between the input stage (source 12) and the output stage (Z L ) of circuit 20, whereas the second planar components 24a, 24b . . . 24 K  are preferably connected by a switch SW 0  as a group to the input stage and to the output stage by means of switch SW K+1 . If desired, the second group of components 24 A , . . . 24 K  could be connected directly to the input stage and to the output stage so as to reduce the number of switches shown in FIG. 3 by eliminating SW 0  and SW K+1 . The switches SW 0 , . . . SW K+1  shown in FIG. 3 are responsive to the switch control module 16 in a similar manner as described with regard to the circuit arrangement 10 in FIG. 2. The circuit arrangement 20 of FIG. 3 has a reduced number of switches (SW) relative to FIG. 2 and accomplishes such a reduction by means of the compact overlapping arrangement of the first and second planar components which may be described with reference to FIGS. 4, 5, 6 and 7. 
     The group of second planar inductive components 24 a , . . . 24 K  (also referred to herein as the &#34;secondaries&#34;) representatively shown as 24 in FIG. 4, comprises a nonconductive substrate 26 on which is mounted a winding 28. In some application the windings are stiff and the substrate 26 is not required. The winding 28 is continuous from a first end 24 a1  to a second end 24 a2 . The reference numbers 24 a1  and 24 a2  are used so as to relate the description of this winding 28 to the description of the impedance elements 24 a , . . . 24 K  of FIG. 3. The winding 28 is preferably, but need not be, in the form of a printed conductor, such as may be formed by conventional screen printing methods. The winding 28 may also be formed by other methods of forming conductors on substrates, such as etching, vacuum deposition as well as forming a conductor in free space (i.e. machined, molded, cast, electroformed, etc.) and the like without departing from the scope of the invention. Thus, the terms &#34;printing&#34; and &#34;forming&#34;  as used herein encompasses and should be understood to include any method of creating a planar conductor. Winding 28 is illustrated as a spiral with a right-angle orientation, but it should be understood that other forms of spirals can be used for winding 28 without departing from the scope of the invention. Likewise, it should be understood that the winding 28 need not be printed, but can be fabricated from a wire, for example, and adhered to the substrate 26 in a suitable manner, such as by an adhesive. 
     The group of first planar inductive components 22a, . . . 22 K  (also referred to herein as the &#34;primaries&#34;) representatively shown as 22 in FIG. 5, is preferably, but not necessarily, fabricated in the same method as the winding 28 and comprises a non-conductive substrate 30 on which is mounted a winding 32. The winding 32 may be formed as well in free space without a substrate. In FIG. 5, a single primary winding 32 is shown as having ends 22 a1  and 22 a2 . If desired, the first planar inductive component 22, as well as the second planar conductive component 24 may comprise multiple windings as shown in FIG. 6. 
     In FIG. 6 showing the second planar component 24, four (4) secondary windings 34, 36, 38 and 40 are illustrated, although it should be understood that the precise number of windings is not critical to the invention. Each of the secondary windings 34, 36, 38 and 40 comprise a pair of conductors, (i.e., γ=2) labelled 42, 44; 46, 48; 50, 52; and 54, 56 respectively. Each of the conductors 42, 44, 46, 48, 50, 52, 54 and 56 is continuous between a first end a (indicated by an arrow) and a second end b (indicated by an arrow). Ends 42a, 44a; 46a, 48a; 50a, 52a; and 54a, 56a are electrically connected in pairs that is, 42 with 44, 46 with 48, 50 with 52, and 54 with 56 as are ends 42b, 44b: 46b, 48b: 50b, 52b and 54b, 56b to define the four secondary windings 34, 36, 38 and 40 respectively. The electrical pairs connections between these windings are illustrated in FIG. 6 by eight (8) jumpers 58. 
     As described with regard to secondary winding 28, conductors 42, 44, 46, 48, 50, 52, 54 and 56 may be printed or otherwise formed on substrate 30. Likewise, secondary windings 34, 36, 38 and 40 preferably have the &#34;same shape&#34; with mirror image symmetry as the primary winding 32. That is, in the illustrated embodiment of FIGS. 4-6, the secondary windings of FIG. 4 are in the shape of a spiral with right-angle orientations while the primary winding 32 of FIG. 5, has mirror image symmetry and thus shaped as a spiral with left hand orientation. 
     Although each secondary windings 34, 36, 38 and 40 is shown in FIG. 6 as a pair of conductors, any number of conductors, as required for a given application, can be used to comprise a secondary winding 34, 36, 38 and 40. Similarly, any number of conductors may be used for the primary winding 32 of FIG. 5. Further, although the primary winding is shown in FIG. 5 as comprising a single winding, any number of conductors to form any number of primary windings, as required by a given application, can be used. Further, preferably, although not necessarily, substrate 26 of FIG. 4 and substrate 30 of FIGS. 5 and 6 are each provided with central openings 60 and 59, respectively, for locating the first planar inductive component 22 and the second planar inductive component 24 relative to one another and also relative to external circuitry. Both the first and second components 22 and 24, respectively, are provided with suitable connectors and associated wiring (not shown) to enable them to be connected to other circuits and to other circuit elements for use as an impedance matching circuit. The precise form of connections is immaterial to the present invention. 
     To form the impedance matching device 20 of the present invention, the first planar inductive component 22 and the second planar inductive component 24 are overlapped and arranged a short distance apart from each other with the primary winding 32 and the secondary winding 28 of FIG. 4 or the secondary windings 34, 36, 38 and 40 of FIG. 6 arranged to face each other. In this configuration, the first and second components are referred to as having mirror-image symmetry. A non-conductive spacer means may be used to maintain a fixed distance between the first and second planar components. The spacer means may consist of a single sheet of non-conductive material which has substantially the same surface area as the first component 22 and the second component 24, or may comprise a plurality of smaller individual spacers located at preselected locations. In this arrangement, the first component 22 and the second component 24 are advantageously coupled magnetically (i.e., inductively) and capacitively. When the first component 22 and the second component 24 are arranged facing each other with mirror-image symmetry, they form an impedance matching device according to the present invention. For convenience, the impedance matching device of the present invention will sometimes be referred to herein as a &#34;P-section&#34;. The term &#34;P-section&#34; is a coined term and is used as short-hand reference to the planar structure (i.e., P for planar) features of the present invention, and has no significance in the RF and microwave art other than such a reference. 
     One embodiment of the P-section may modelled as illustrated in FIG. 7. The first planar inductive component 22, having first (22 a1 ) and second (22 a2 ) ends previously described with reference to FIG. 5, may be envisioned as a primary winding of a transformer, whereas the second planar inductive component 24, having first (24 a1 ) and second (24 a2 ) ends previously described with reference to FIG. 4, may be envisioned as the secondary winding of the same transformer. For the embodiment illustrated in FIG. 7, the first and the second planar inductive components are illustrated to have substantially the same number of windings, however, it should be recognized that the turns ratio for the first and second planar components may be a number greater than 1. In all such embodiments, since the first planar inductive component 22 and the second planar inductive component 24 face each other, there is both a capacitive C c  and inductive coupling as shown schematically in FIG. 7, between the primary and secondary windings shown schematically as a transformer as well as a capacitor. For ease of reference, the primary terminals are labelled P1 and P2 and the secondary terminals are labelled 1A and 1B. The secondary terminals 1A and 1B are shown to have connected a switch SW across them. The operation of the switch (SW) effects both the primary and secondary windings and is of primary importance of the present invention, and may be further described with reference to FIG. 3. 
     The switches SW 1 , . . . SW K  are an essential part of the impedance matching device 20 of FIG. 3 and have respective overlapping windings 22 a , . . . 22 K  and 24 a , . . . 24 K  that provide in effect as many transmission variable segments as compared to the transmission segments 14 1 , . . . 14 N  of FIG. 2 yet while saving about half of the number of switching. These overlapping sections of FIG. 3 are provided within smaller space compared to the arrangement of FIG. 2. The operation of switches SW 1 , . . . SW K , along with the overlapping windings provides for a reduction in the number of related switches SW 1 , . . . SW K  shown in FIG. 3 relative to those switches shown in FIG. 2. The number of switches (M) needed for the operation of the impedance matching device 20 of FIG. 3, relative to the number of switches needed for the impedance matching device 10 of FIG. 2, may be expressed as: 
     
         M=N/γ+2                                              (3) 
    
     where N is the original number of switches shown in FIG. 2 and γ is the number of overlapping &#34;loops&#34; (see expression 1). The quantity 2 in expression (3) corresponds to the preferred switches SW 0  and SW K+1  shown in FIG. 3. 
     The switch count reduction realized by the circuit arrangement of FIG. 3 because in operation the current I 1  flowing in the primary windings 22a, . . . 22 K  of circuit 20 produces a variable magnetic field per Ampere&#39;s Law. The resulting magnetic field φ 1  produces a voltage across the corresponding secondary winding 24 a , . . . 24 K  generally indicated in FIG. 3. When the corresponding secondary winding 24 a , . . . 24 K  is shorted out by the operation of its corresponding switch, e.g., SW 1  shorting out 22 a  both components also generally indicated in FIG. 3, a corresponding current I 2  is produced on the secondary winding due to the presence of the magnetic field φ 1 . This current I 2  creates a secondary magnetic field φ 2  that cancels out φ 1  and therefore acts as if a short was placed immediately across the corresponding primary winding 22 a , . . . 22 K . Thus, a single switch such as SW 1 , or SW K  can short out both the secondary winding 24 a , or . . . 24 K  and the respective primary winding 22 a , or . . . 22 K . 
     The operation of a single switch shorting out the secondary winding and effectively shorting out the primary winding allows for broad band frequency operation. At the lower end of the frequency range, the primary windings 22 a  . . . 22 K  act as a variable length transmission line and can typically perform over two octaves, wherein each octave is defined as a region between a given frequency f and either twice that frequency (2f) or half that frequency (f/2). Thus, two octaves may be expressed as band of frequencies of f/2 to 2f. 
     In practice, the coupling between the primary windings 22 a , . . . 22 K  and 24 a , . . . 24 K  is not perfect and as a result the primary windings 22 a , . . . 22 K  experience a parallel resonant condition, acting as an open circuit between the generator 12 and the load Z L , at the higher frequencies of the desired performance band. This parallel resonance is equivalent to a high impedance being imposed between the source 12 and the load Z L . This parallel resonance hinders the desired impedance matching in an original configuration in which the primary windings were not directly connected to the load (Z L ). However, it was recognized during the development of this invention that the length of the secondary windings 24 a  . . . 24 K  can still be adjusted regardless of the parallel resonant condition of the primary windings 22 a  . . . 22 K . Thus, in the upper band of the desired frequency range, the secondary windings 24 a  . . . 24 K  perform the impedance matching covering an additional two octaves in the upper band of the frequencies. The obtainable overall frequency band of the impedance matching device 20 may be from f/4 to 4f wherein the primary winding 22 a  . . . 22 K  supplies the desired characteristic impedance in the lower frequency band from f/4 to f, and the secondary winding 24 a  . . . 24 K  supplies the desired characteristic impedance in the upper frequency band from f to 4f. 
     The frequency f is commonly selected to correspond to the frequency of the source 12, which typically has values from 2-32 MHz. However, the frequency f for the device 20 of the present invention may be selected to accommodate any impedance matching function desired for the present invention so as to yield a desired frequency band between four octaves of any selected frequency f. Thus, a broad frequency coverage is obtained by the present invention having a minimum number of electronic switches (see expression (3)), and a compact circuit arrangement (overlapping primary 22 a , . . . 22 K  and secondary 24 a , . . . 24 K  windings), as described with regard to impedance matching device 20 of FIGS. 3-7. 
     In addition to providing the reduced number of switches, along with an improved frequency band characteristic, the P-section impedance matching device 20 of the present invention reduces or even eliminates the problems previously discussed in the &#34;Background&#34; with regard to the π and T matching circuits. As discussed, the and T sections each has an inductor that may be shorted out for short durations corresponding to some one or more frequencies within the band of frequencies supplied by the source 12 to the load Z L  antenna. This shorting of the inductor may be needed so that π or T section provides the proper impedance matching characteristic to the antenna. This high speed shorting generates a high current, e.g., 20 amperes, along with a high voltage, e.g., 10 kV, which create high stress conditions that may cause premature failures of the related solid state switching devices. The device 20 of FIG. 3 (P-section) of the present invention utilizes a magnetic field between the first (22 a , . . . 22 K ) and second (24 a , . . . 24 K ) planar inductive components to provide its impedance matching so as to transform the load impedance (Z L ) into the desired characteristic (Z 0 ). This magnetic field technique does not create the high stress conditions caused by the π or T matching circuits. 
     The broad frequency response (f/4 to 4f) of the present invention is obtained by the impedance matching device 61 shown in FIG. 8 as comprising a plurality of sets of first and second planar inductive components which are 62 a , 62 b  ; 64 a , 64 b  ; and 66 aK , 66 bK . The first components 62 a , 64 a  and 66 aK  are similar to the first components 22 a  . . . 22 K  described with reference to FIG. 3, whereas the second components 62 b , 64 b  and 66 bK  are similar to 24 a  . . . 24 K  also of FIG. 3. The first (primary) components 62 a , 64 a , . . . 66 aK  may be comprised of multiple windings as indicated by the break-lines, shown in FIG. 8, for each set 62 a , 64 a , . . . 66 aK . Each of the secondary windings 62 b , 64 b , . . . 66 bK  has a corresponding switching device SW 1 , SW 2 , SW K  which operates in a manner as previously described with reference to FIG. 3. The switches SW 1 , SW 2 , . . . SW K  are shown as having respective terminals A and B related to the effective shorting of secondary windings 62 b , 64 b  and 66 bK . These first and second planar inductive components are segmented and separated from each other into any predetermined number of sets with each set having a predetermined impedance characteristic which is selectable to provide a segment of the overall predetermined impedance over the desired frequency range of operation. The predetermined segment is further subdivided into selectable impedances provided by each of the primary windings 62 a , . . . 66 aK  and each of the secondary windings 62 b , . . . 66 bK . This further subdivided feature is equally applicable to the impedance matching device 20 of FIG. 3 along with other embodiments of the invention to be described. 
     The arrangement of FIG. 8 is similar to FIG. 3 in that the primary windings 62 a , 64 a  and 66 aK  are serially connected between the input stage (source 12) and the output stage (Z L ). Similarly, the group of secondary primary windings 62 b , 64 b , . . . 66 bK  are preferably connected to the input stage by means of switch SW 0  and to the output stage by switch SW K+1 . The primary windings 62 a , 64 a , . . . 66 aK  and their respective secondary windings 62 b , 64 b  and 66 bK  operate in the same manner, in response to the corresponding switching means SW 1  . . . SW K , as discussed with regard to FIG. 3. 
     A further impedance matching device 70 of the present invention is shown in FIG. 9. The impedance matching device 70 is similar to the previously discussed device 60 with the exception that the sets of the first 62 a , 64 a , 66 aK  and second planar 62 b , 64 b , . . . 66 bK  inductive components are compressed in size compared to FIG. 8 with each set having a closed contour shape. FIG. 9 uses the broken line symbol to indicate that the each of the loops of the first planar component 62 a , 64 a  and 66 aK  has γ turns which is defined by the previously discussed expression (1). FIG. 9 also illustrates SW 0 , SW 1 , SW 2 , SW K  and SW K+1 . For the sake of clarity, the switches SW 1 , SW 2 , . . . SW K  having their respective illustrated terminals 1A-1B, 2A-2B, and KA-KB are shown away from their related windings 62 b , 64 b , . . . 66 bK . Further the switch SW 0  is shown as being connected between the input (IN) and terminal 1A whereas switch KW K+1  is shown as being connected between the output (OUT) and terminal KB. The set 66 aK  and 66 bK  forms an inner closed contour with the sets 64 a , 64 b  and 62 a , 62 b  having corresponding increasing dimensions to provide increasing closed contour shapes in a fanned-out arrangement. 
     A still further embodiment of the present invention is shown in FIG. 10 for a impedance matching device 80. The device 80 comprises multiple circuit arrangements 70, discussed with regard to FIG. 9, that are connected in cascade and arranged inside of an enclosure 82 which is grounded. Such a planar transmission line enclosed in a grounded enclosure is known as a stripline in the RF microwave art. A stripline that is suspended in air, without the &#34;substantial&#34; use of dielectric material, is also known as a suspended stripline. Thus, device 80 is denoted as multilayer magnetically coupled suspended stripline (MMCSS). 
     It should now be appreciated that the practice of the present invention provides for an impedance matching device having various embodiments each of which reduces the number of switching elements needed, reduces the stress conditions of the remaining switching elements, and provides for transforming load Z L  into a desired characteristic impedance Z 0  over a wide range of high power and high frequency operation. 
     The present invention may be embodied in other specific forms without departing from the spirit or essential attributes thereof and, accordingly, reference should be made to the appended claims, rather than to the foregoing specification, as indicating the scope of the invention.