Abstract:
A control circuit for causing a phase lock loop (PLL) frequency synthesizer to achieve a fast phase lock time while also providing improved loop performance during normal phase locked operation. The phase locking time of the PLL is minimized by initially configuring the PLL to operate in a fractional mode with high frequency signals presented to the inputs of the loop phase detector, thereby producing a fast phase lock time. Once the PLL has achieved phase lock, its operation mode is transitioned to either an integer mode or an open loop mode without loss of phase lock, thus causing lower frequency signals or no signals, respectively, to be presented to the inputs of the loop phase detector and thereby significantly reducing spurious signal tones.

Description:
FIELD OF THE INVENTION 
     The present invention relates to Phase Locked Loops (“PLL&#39;s”), and in particular, to a PLL with a fast lock time. 
     BACKGROUND OF THE INVENTION 
     Phase Locked Loops (“PLL&#39;s”) are systems which allow different signals in different systems to track with one another. One application of a PLL is in digital communication systems. In digital communication systems, the receiving system must be able generate the various frequencies necessary for processing various received signals. For example, a receiving system must be able to synthesize a specific frequency for mixing down the received signals. To accomplish this, a reference frequency is applied to the input of a PLL and a system division ratio of the PLL is set so that the output is some scaled up factor of the input. One problem common to all PLL&#39;s is that the output frequency of the PLL system will require a certain amount of time to lock up to a given input frequency. This is known as the lock time. The lock time of a PLL is highly non-linear and very difficult to control. It is desirable to reduce the lock time as much as possible so as to reduce the amount of time the system must wait for the PLL to lock. 
     Frequency synthesis using a PLL is well known in the art. One example of a prior art PLL frequency synthesizer is shown in FIG.  1 . The PLL  100  of FIG. 1 includes a phase detector (“PD”)  110 , a loop filter  120 , a voltage controlled oscillator (“VCO”)  130 , a reference divider  101  having a divider ratio of R, and a feedback divider  102  having a divider ratio of B. The PLL  100  of FIG. 1 is known as an Integer Divider because the frequency at the output is an integer multiple of the frequency at the input of the phase detector. A fixed reference signal Fref is transmitted to the reference divider  101  and then to one input of the Phase Detector. The output of the VCO is divided by the feedback divider and input to the other input of the Phase Detector. Assuming the system is locked the following equation is satisfied: 
     
       
         F1=F2 
       
     
     
       
         F1=Fref/R 
       
     
     
       
         F2=Fout/B 
       
     
     and 
     
       
         Fout=Fref (B/R) 
       
     
     By way of example, if Fref=10 Mhz, R=100, and B=5, then 
     
       
         Fout=500 kHz 
       
     
     Thus it can be seen that Fout will be some integer fraction of the reference frequency Fref. 
     Another example of a prior art PLL frequency synthesizer is shown in FIG.  2 . The PLL  200  of FIG. 2 includes a phase detector (“PD”)  210 , a loop filter  220 , a voltage controlled oscillator (“VCO”)  230 , a reference divider  201  having a divider ratio of R, a feedback divider  202  having a divider ratio of B, and a prescaler divider  203  having a divider ratio of K. The PLL  200  of FIG. 2 is known as an Integer Divider with Prescaling. A fixed reference signal Fref is transmitted to the reference divider  201  and then to one input of the phase detector. The output of the VCO is divided by the prescaler divider and the feedback divider, and applied to the other input of the phase detector. Again, assuming the system is locked the following equation is satisfied: 
     
       
         F1=F2 
       
     
     
       
         F1=Fref/R 
       
     
     
       
         F2=F3/B 
       
     
     
       
         F3=Fout/K 
       
     
     and 
     
       
         Fout=Fref*K*(B/R) 
       
     
     Therefore, with prescaling, if Fref=10 Mhz, R=100, B=5, K=10, then 
     
       
         Fout=5 MHz 
       
     
     Thus it can be seen that Fout will be some integer fraction of the reference frequency Fref multiplied by the prescaler value. 
     FIG. 3 shows another example of a prior art PLL used for frequency synthesis. The PLL  300  of FIG. 3 includes a phase detector  340 , a loop filter  350 , a VCO  360 , a reference divider  310  having a divider ratio of R, a feedback divider  320  having a divider ratio of B, an auxiliary divider  325  having a divider ratio of A, and a dual modulus prescaler  330  which can be configured to have a divide ratio of either K or K+1. Again the reference frequency is divided down before being applied to the input of the phase detector. The output signal is fed back through the dual modulus prescaler which feeds a signal to both the feedback divider and the auxiliary divider. The output of the feedback divider is applied to the other input of the phase detector. 
     To understand the operation of the PLL  300  of FIG. 3 by way of example, assume that both the feedback divider and auxiliary divider are DOWN counters, referred to here as B-counter and A-counter respectively. The output of the B-counter is transmitted to the input of the phase detector, and additionally, over LOAD line  301  to the load inputs of both the A-counter and B-counter. Therefore, every time the B-counter counts to zero and outputs a pulse, it will reset both the B-counter and the A-counter to their initial values. The dual modulus prescaler  330  is a divider which can divide the output, Fout, by two different integer values (in this case K and K+1) in accordance with the Prescaler Control line  302  from the A-counter. Assuming the system is locked and the B-counter has just counted down to zero and output a pulse to an input of the phase detector as well as reset the B-counter and A-counter, the signal Fout at the VCO output will be received by the dual modulus prescaler. Initially, the prescaler will divide the VCO output, Fout, by K+1 and the prescaler output will begin to supply pulses to both the B-counter and A-counter, causing each to begin to count down. When the A-counter reaches zero, a signal is transmitted over the Prescaler Control line which causes the dual modulus prescaler to reconfigure itself to stop dividing by K+1 and begin dividing by K. Thereafter, the prescaler will divide the output, Fout, by K and the prescaler output will cause the B-counter to continue to count down until it reaches zero. When the B-counter reaches zero, it outputs another pulse to the input of the phase detector. This pulse also causes the A-counter and B-counter to reset. Therefore, it can be seen that for every pulse, Npd, at the input of the phase detector, there will be Ntot pulses at the output of the VCO. Ntot can be determined by noting that while the A-counter is counting down the prescaler is dividing by K+1. Therefore, the total number of pulses at the VCO output required for the A-counter to count down to zero is A(K+1). Thereafter, the prescaler divides by K, so the total number of pulses at the VCO output required for the B-counter to finish its count down to zero is (B−A)(K) (note: the B-counter and A-counter were counting down together). Therefore, the total number of pulses at the output of the VCO, Ntot, is given by: 
     
       
         Ntot=A(K+1)+(B−A) K For one pulse, Npd, into the phase detector. 
       
     
     
       
         Ntot=A+BK 
       
     
     or in terms of the period, 
     
       
         Tpd=Tout (A+BK) 
       
     
     
       
         Tpd=1/F1 and Tout=1/Fout 
       
     
     
       
         Fout=F1 (A+BK) 
       
     
     Therefore, the following equations are satisfied: 
     
       
         F1=Fref/R=F2=F3/B 
       
     
     
       
         F3=Fout/(K+1) For A cycles (i.e. while A is counting down) 
       
     
     
       
         F3=Fout/K For B−A cycles (i.e. while B is counting down after A has finished counting down. 
       
     
     and the VCO output frequency is given by: 
     
       
         Fout=(Fref/R)*(A+BK) 
       
     
     It can be seen that a necessary condition of this system is that the B-counter must contain a value which is equal to or larger than the value contained within the A-counter. It can be seen that other implementations besides DOWN counters could be used to implement the system of FIG.  3 . Therefore, a more generic condition for the system is that the auxiliary divider must signal the prescaler and become inactive before the B-divider. This type of PLL frequency synthesizer is called a dual modulus prescaler integer PLL. 
     In many systems it is advantageous to synthesize a frequency which is a non-integer multiple or fraction of a reference frequency. Such frequency synthesizers are called fractional frequency synthesizers and achieve faster phase lock since the reference frequency can be increased. An example of a prior art PLL used as a fractional frequency synthesizer is shown in FIG.  4 A. The fractional frequency synthesizer PLL  400  of FIG. 4 includes a phase detector  450 , a loop filter  460 , a VCO  470 , a reference divider  410  having a divider ratio of R, a feedback divider  420  having a divider ratio of N, a pulse swallowing circuit  430 , an accumulator (“accumulator”)  440 , an N-register  425  for storing the integer portion of a system divisor number, and an F-register  445  for storing the fractional portion of a system divisor number. The system division ratio of such a fractional PLL system is N.F, where N is the integer part and F is the fractional part. In other words, Fout=Fref (N.F). The integer and fractional parts of the division ratio are stored in the N-register and F-register, respectively. 
     To illustrate the operation of the fractional PLL  400 , assume the system is locked and that the desired division ratio of the system is: 
     
       
         N=5 F=3 and N.F=5.3 
       
     
     
       
         Fout=Fref (5.3) for R=1 
       
     
     Therefore, for every  10  cycles of Fref, there will be  53  cycles of Fout. FIG. 4B illustrates the signal F1=Fref (R=1) at the input of the phase detector as well as the contents of the accumulator accumulator. During the first cycle of Fref, referred to here as the first reference cycle, the PLL attempts to divide the output of the VCO, Fout, by N.F=5.3, but this it cannot do. Instead, during the first cycle, the system divides Fout by the integer portion of the fractional divisor, N=5, which is loaded into the feedback divider. Therefore, during the first reference cycle, there is an error between F1 and F2 equal to 0.F * Fout 0.3 * Fout. The error in Fout is going to show up as a phase error in F2 at the input of the phase detector. This phase error can be represented and accounted for by using the accumulator to keep track of the error in Fout. This is accomplished by loading the value of 0.F into the accumulator and using F1 (=Fref in this case) to accumulate the error in each reference cycle. This is shown in FIG.  4 B. During the first reference cycle, 0.F=0.3 is loaded into the accumulator. During the next cycle, an additional error of 0.F=0.3 is added to the current error. This continues until the phase error between F1=Fref and F2=Fout/N becomes greater than one full cycle of Fout=N*F2 (i.e., 2π radians of Fout). This corresponds to the point where the fractional error in accumulator becomes greater than unity. As shown in FIGS. 4A and 4B, when the fractional error in the accumulator exceeds unity, an overflow signal (“OVF”) in the accumulator signals the pulse swallowing circuit to remove a pulse from the feedback path. The result of removing the pulse is that the feedback divider will not register one pulse of Fout. This is the same as if the feedback divider had divided by N+1=6, rather than by N=5 during that reference cycle, which will essentially delay F2 and reduce the phase error between F1 and F2. A residual error of 0.2 is maintained in the accumulator as shown in FIG.  4 B. which represents the phase error between F1 and F2 after the pulse is swallowed. The error will again accumulate with each reference cycle. This process will proceed across  10  reference cycles as shown in FIG.  4 B. Note that the accumulator overflows three times during the ten reference cycles. Therefore, over ten reference cycles the VCO has put out 10 * 5 cycles plus three additional pulses which were not transmitted to the feedback divider. In other words, 10 cycles of Fref produced 53 cycles of Fout, or 
     
       
         10*Tref=53*Tout 
       
     
     
       
         10/Fref=53/Fout 
       
     
     
       
         Fout=5.3 Fref 
       
     
     
       
         Fout=N.F Fref 
       
     
     which is what was desired. 
     All of the above mentioned PLL frequency synthesizer architectures share problems associated with traditional PLL&#39;s. One problem is that at low reference frequencies, the PLL&#39;s require a longer time to lock than at high frequencies. For fractional frequency synthesizers, another problem is that at high operating frequencies PLL&#39;s have high levels of spurious tone power caused by the averaging and which can reduce system performance. 
     Accordingly, it would be desirable to have a PLL which could achieve a fast lock time without the spurious tone power resulting from high frequency operation. 
     SUMMARY OF THE INVENTION 
     The present invention provides a phase locked loop frequency synthesizer with a fast lock time that provides superior performance during normal operation. Lock time of the PLL is reduced by initially configuring the PLL in a fractional mode with a high frequency signal at the input of the phase detector. With a high frequency signal at the input of the phase detector, the PLL will achieve a fast lock time. After the PLL is locked, the system smoothly transitions into an integer mode, or alternatively into an open loop mode, without loosing lock. When configured in integer mode, the frequency of the signal at the input of the phase detector is reduced, thereby eliminating the averaging, thus reducing spurious tone power and resulting in improved performance. When configured in an open loop mode, the loop is “opened” by causing the output terminal of the phase detection circuit (e.g., a charge pump output) to enter a high impedance state of operation while preventing leakage of charge from such output terminal normally caused by the output stage of the phase detection circuit, thereby eliminating spurious tone power and resulting in improved performance. 
     In accordance with one embodiment of the present invention, a dual mode control circuit for a phase lock loop (PLL) includes a reference signal frequency divider circuit, a feedback signal frequency divider circuit and a control circuit. The reference signal frequency divider circuit is configured to couple to a PLL and receive at least one reference divider control signal and in accordance therewith receive and frequency divide an input reference signal having an input reference signal frequency and in accordance therewith provide one or more divided reference signals having one or more divided reference signal frequencies and phases. The feedback signal frequency divider circuit is configured to couple to the PLL and receive at least one feedback divider control signal and in accordance therewith receive and frequency divide a PLL feedback signal having a PLL feedback signal frequency and in accordance therewith provide one or more divided feedback signals having one or more divided feedback signal frequencies and phases. The control circuit, coupled to the reference and feedback signal frequency divider circuits, is configured to couple to the PLL, receive a PLL lock signal from the PLL and receive and process one of the one or more divided feedback signals and in accordance therewith provide the at least one reference divider control signal and the at least one feedback divider control signal. The control circuit, in accordance with the PLL lock signal, and the reference and feedback signal frequency divider circuits, in accordance with the at least one reference divider control signal and the at least one feedback divider control signal, transition between first and second circuit operation modes when: the PLL lock signal indicates that the PLL has transitioned between unlocked and phase locked states of operation; and the processing of the one of the one or more divided feedback signals indicates that a phase difference between the one of the one or more divided feedback signal phases and a desired signal phase transitions between outside and inside of a predetermined phase difference range. 
     In accordance with another embodiment of the present invention, a method of controlling a phase lock loop (PLL) in accordance with dual PLL operation modes includes the steps of: 
     coupling to a PLL; 
     receiving from the PLL a PLL feedback signal having a PLL feedback signal frequency; 
     receiving at least one reference divider control signal and in accordance therewith receiving and frequency dividing an input reference signal having an input reference signal frequency and in accordance therewith generating one or more divided reference signals having one or more divided reference signal frequencies and phases; 
     receiving at least one feedback divider control signal and in accordance therewith frequency dividing the PLL feedback signal and in accordance therewith generating one or more divided feedback signals having one or more divided feedback signal frequencies and phases; 
     receiving a PLL lock signal from the PLL and receiving and processing one of the one or more divided feedback signals and in accordance therewith generating the at least one reference divider control signal and the at least one feedback divider control signal; and 
     transitioning between first and second operation modes in accordance with the PLL lock signal, the at least one reference divider control signal and the at least one feedback divider control signal when 
     the PLL lock signal indicates that the PLL has transitioned between unlocked and phase locked states of operation, and 
     the processing of the one of the one or more divided feedback signals indicates that a phase difference between the one of the one or more divided feedback signal phases and a desired signal phase transitions between outside and inside of a predetermined phase difference range. 
    
    
     These and other features and advantages of the present invention will be understood upon consideration of the following detailed description of the invention and the accompanying drawings. 
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The present invention is illustrated by way of example and not limitation in the figures of the accompanying drawings, in which like references indicate similar elements, and in which: 
     FIG. 1 is a diagram of an integer divider PLL frequency synthesizer as known in the prior art. 
     FIG. 2 is a diagram of an PLL integer divider with prescaling as known in the prior art. 
     FIG. 3 is a diagram of a conventional dual modulus prescaler integer PLL. 
     FIG. 4A is a diagram of a conventional fractional frequency synthesizer PLL as known in the prior art. 
     FIG. 4B is a waveform diagram illustrating the operation of the fractional frequency synthesizer PLL of FIG.  4 A. 
     FIG. 5 is a diagram of a fast locking dual mode PLL according to one embodiment of the present invention. 
     FIG. 6A is a diagram of a fast locking PLL configured in fractional mode according to one embodiment of the present invention. 
     FIG. 6B is a waveform diagram illustrating the fractional mode operation of the fast locking PLL of FIG. 6A according to one embodiment of the present invention. 
     FIG. 7A is a waveform diagram illustrating the fractional mode operation of the fast locking PLL of FIG. 6A according to another embodiment of the present invention. 
     FIG. 7B is a waveform diagram illustrating the actual phase error of the fast locking PLL of FIG. 6A according to one embodiment of the present invention. 
     FIG. 8 is a diagram of a fast locking PLL configured in integer mode according to one embodiment of the present invention. 
     FIG. 9 is a diagram of a dual mode fast locking PLL according to one embodiment of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Referring to FIG. 5, a PLL  500  according to one embodiment of the present invention is shown. PLL  500  includes a reference divider  510 , a feedback divider  520  which can be configured in either a fractional mode or an integer mode, a phase detector  530 , a loop filter  540 , a voltage controlled oscillator  550 , and a phase error tracking system  560 . Initially, the PLL  500  is configured in a high frequency fractional mode. In high frequency fractional mode, there is a higher frequency of pulses at the input of the phase detector that will result in a fast lock time. After the loop has locked, the phase error tracking system  560  monitors the changes in phase error for each pulse cycle at the input to the phase detector. When the phase error tracking system detects a pulse cycle with a minimum phase error, the phase error tracking system reconfigures the feedback divider and the reference divider into a low frequency integer mode of operation. In a low frequency integer mode of operation, with minimum or no phase error at the input of the phase detector, the system experiences a reduction in spurious tone power. In one embodiment, when the phase error tracking system reconfigures the system, the gain of the phase detector is also reduced so that the phase detector does not overshoot the frequency of interest. In another embodiment, when the phase error tracking system reconfigures the system, the phase detector output is put into a high impedance state, and the PLL system is configured in a free running mode. 
     FIG. 6A illustrates one embodiment of the present invention when the PLL system is in the initial fractional configuration mode of operation. In fractional mode, PLL  600  includes a reference divider  610  having a divider ratio of R, a phase detector  620 , a loop filter  630 , a voltage controlled oscillator (“VCO”)  640 , a feedback divider  650  having a divider ratio of B, an auxiliary divider  651  having a divider ratio of A, a cycle slip controller  652 , a prescaler  665 , an accumulator (“accumulator”)  670 , a modulus (“modulus”) circuit  671 , and a comparator circuit  680 . In one embodiment, the fractional portion of the system division ratio 0.F is represented as a fraction with the numerator and denominator stored in some form of memory. For the PLL of FIG. 6A the numerator is stored in numerator register  675  and the denominator in denominator register  676 , for example. PLL  600  also includes memory for storing a predetermined exit state. In one embodiment the exit state is stored in an exit state register  685 . 
     The reference frequency, Fref, is supplied to the reference divider, divided by ratio R, and transmitted to the one input of the phase detector at F1. The output of the PLL, Fout, is divided by either N or N+1 in the prescaler  665  and fed back through the feedback divider  650  to the second input of the phase detector at F2. Assuming that the system is locked and that the feedback divider has just transmitted a pulse, the feedback divider and the auxiliary divider have just been re-loaded with their divisor ratios B and A, respectively. Initially, the prescaler control line  653  from the auxiliary divider will signal the prescaler to divide the VCO output, Fout, by N+1. The prescaler will divide the output, Fout, by N+1 for A(N+1) VCO output pulses (i.e., A pulses of the prescaler output). After A(N+1) cycles of Fout, the auxiliary divider  651  signals the prescaler  665  over the prescaler control line  653  to reconfigure itself to divide by N. The prescaler  665  will thereafter divide the VCO output, Fout, by N. After (B−A)(N) pulses of Fout (i.e., B−A pulses of the prescaler output), the feedback divider  650  transmits a pulse to the phase detector  620 . Therefore, over the first reference pulse cycle (i.e., feedback divider output cycle), the frequency of the pulses at F2 is the same as for a dual modulus fractional divider and given by: 
     
       
         F2=Fout/(BN+A) 
       
     
     However, for fractional division, at the end of the first reference pulse cycle there will be a phase error between F1 and F2 as previously described. This error corresponds to the value O.F of the fractional portion of the PLL system division ratio. In the PLL of FIG. 6A, the fractional portion of the system division ratio is represented as a fraction having a numerator and denominator. The numerator value is stored in numerator register  675  and the denominator is stored in denominator register  676 . Both of these values can be set during a system initialization. The numerator register  675  is connected to the A input of the accumulator  670  and the denominator register  676  is connected to the B input of modulus circuit  671 . The accumulator  670  is triggered by the feedback divider  650  during each reference pulse cycle. (A reference pulse cycle, “reference cycle”, or “system cycle”, is the time between each feedback divider pulse, F2, to the phase detector. It may also be referred to as a phase detector cycle.) Therefore, during the first reference cycle the feedback divider triggers the accumulator to load the value of the numerator into the A-input of the accumulator. A modulus operation is then performed on the output of the accumulator and the value in denominator register  676 . The result of the modulus operation is fed back to the B-input of the accumulator and added to the numerator when triggered by the feedback divider at the end of each system cycle. Therefore, for each phase detector pulse, the fractional error is stored and accumulated in the accumulator. This will continue until the fractional error between the pulses becomes greater than the pulse width. This corresponds to the situation where the modulus overflows. If the modulus operation carried out during the given reference cycle results in an overflow (“OVF”), the OVF line  672  signals the cycle slip controller  652  to add an extra N+1 divide during the current reference cycle. This extra N+1 divide will occur after the auxiliary divider  651  has finished dividing by N+1 and has signaled the prescaler  665  to begin dividing by N. In other words, if the result of a modulus operation during a given system cycle results in an OVF signal, the cycle slip controller  652  will signal the prescaler  665  to divide by N+1 for one additional period of the prescaler output (i.e. the prescaler  665  will divide by N+1 for A+1 prescaler output periods). Therefore, during a reference cycle with an OVF condition, the total number of cycles of Fout will be given by: 
     
       
         Ntot=(A+1)(N+1)+(B−A−1) N for one pulse, Npd, into the phase detector. 
       
     
     
       
         Ntot=BN+A+1 
       
     
     
       
         Fout=(Fref/R)*(BN+A+1) 
       
     
     In other words, the VCO output frequency is given by Fout=(Fref/R)*(BN+A) when an overflow event has not occurred for a given system cycle, and the VCO output frequency is given by Fout=(Fref/R)*(BN+A+1) when an overflow event has occurred for a given system cycle. (Notwithstanding these apparent differences in the VCO output frequency Fout, averaging of the VCO control signal by the loop filter causes the frequency to remain constant, albeit with a linearly varying phase.) Table 1 illustrates the accumulator values and the modulus results for a fractional divisor value of F=2/5. 
     
       
         
               
               
               
               
               
             
           
               
                 TABLE 1 
               
               
                   
               
               
                 ACCU A 
                 ACCU B 
                 ACCU OUT 
                 (ACCU) MOD (DENOM) 
                 OVF 
               
               
                   
               
             
             
               
                 2 
                 0 
                 2 
                 2/5 = 0r2 
                 0 
               
               
                 2 
                 2 
                 4 
                 4/5 = 0r4 
                 0 
               
               
                 2 
                 4 
                 6 
                 6/5 = 1r1 
                 1 
               
               
                 2 
                 1 
                 3 
                 315 = 0r3 
                 0 
               
               
                 2 
                 3 
                 5 
                 5/5 = 1r0 
                 1 
               
               
                 2 
                 0 
                 2 
                 2/5 = 0r2 
                 0 
               
               
                   
               
             
          
         
       
     
     The operation of the PLL  600  is also illustrated FIG.  6 B. During the first reference cycle, the value stored in the numerator register is added to the output of the modulus circuit. Initially, the value of the modulus circuit is zero. Therefore, for the example shown in Table 1, the value of two is added to zero and the output of the modulus circuit becomes two. In the second reference cycle, the value of two in the numerator register is added to the value of two at the output of the modulus circuit. The result at the accumulator output (four) undergoes a modulus operation with the value in the denominator register (five). The result (four) is again fed back to the input of the accumulator for the next reference cycle. During the third reference cycle the value in the numerator register (two) is added to the previous modulus result (four) and the accumulator result (six) undergoes a modulus operation with the value in the denominator register (five). This time an overflow occurs as shown in FIG.  6 B. The modulus overflow is transmitted at the OVF output of the modulus circuit to the cycle slip controller  652 . The cycle slip controller signals the prescaler to include an extra N+1 divide in the current reference cycle. As shown in Table 1 and in FIG. 6B, there are two such overflow events every five reference cycles. Therefore, the frequency at the output of the PLL  600  is given by: 
     
       
         Fout=(Fref)*(BN+A) for 3 out of 5 cycles 
       
     
     
       
         Fout=(Fref/R)*(BN+A+1) for 2 out of 5 cycles 
       
     
     
       
         Fout=(3[(Fref/R)*(BN+A)]+2[(Fref/R)*(BN+A+1)])5/ 
       
     
     
       
         Fout=(Fref/R)*(BN+A)+(Fref/R)*(2/5) 
       
     
     By way of example in a typical GSM communication system, given the desired output frequency Fout=1000.4 MHz and Fref=13 MHz, by letting N=16 (N+1=17), A=8, B=62, and R=13 we get: 
     
       
         N.F=Fout/(Fref/R)=1000.4 MHz/(13 MHz/13)=1000.4=1000+2/5 
       
     
     
       
         N=1000 and F=2/5 
       
     
     
       
         Fout=(13 MHz/13)*((62)*(16)+(8))+2/5(13 MHz/13) 
       
     
     
       
         Fout=1000.4 MHz 
       
     
     as desired. 
     FIG. 7A illustrates another embodiment of the present invention. In the embodiment corresponding to the waveforms of FIG. 7A, the system reference frequency Fref is 13 MHz and the desired system output frequency Fout is 1001.4 MHz. Furthermore, in this embodiment the fractional divisor N.F is 
     
       
         N.F=385 2/13 
       
     
     This system divisor ratio is achieved by letting N=16 (N+1=17), A=1, B=24, R=5, and F=N/D=2/13. Therefore, the system output frequency is given by: 
      Fout_ave=(Fref/R)*(BN+A)+(Fref/R)*(N/D) 
     
       
         Fout_ave=(13 MHz/5)*((24)(16)+1 )+(13 MHz/5)*(2/13) 
       
     
     
       
         Fout_ave=1001 MHz+0.4 Mhz=1001.4 MHz 
       
     
     Table 2 illustrates the accumulator values and the modulus results for a fractional divisor value of F=2/13. 
     
       
         
               
               
               
               
               
             
               
               
               
               
               
             
           
               
                 TABLE 2 
               
               
                   
               
               
                 ACCU A 
                 ACCU B 
                 ACCU OUT 
                 (ACCU) MOD (DENOM) 
                 OVF 
               
               
                   
               
             
             
               
                   
               
             
          
           
               
                 2 
                 0 
                 2 
                  2/13 = 0r2 
                 0 
               
               
                 2 
                 2 
                 4 
                  4/13 = 0r4 
                 0 
               
               
                 2 
                 4 
                 6 
                  6/13 = 0r6 
                 0 
               
               
                 2 
                 6 
                 8 
                  8/13 = 0r8 
                 0 
               
               
                 2 
                 8 
                 10 
                 10/13 = 0r10 
                 0 
               
               
                 2 
                 10 
                 12 
                 12/13 = 0r12 
                 0 
               
               
                 2 
                 12 
                 14 
                 14/13 = 1r1 
                 1 
               
               
                 2 
                 1 
                 3 
                  3/13 = 0r3 
                 0 
               
               
                 2 
                 3 
                 5 
                  5/13 = 0r5 
                 0 
               
               
                 2 
                 5 
                 7 
                  7/13 = 0r7 
                 0 
               
               
                 2 
                 7 
                 9 
                  9/13 = 0r9 
                 0 
               
               
                 2 
                 9 
                 11 
                 11/13 = 0r11 
                 0 
               
               
                 2 
                 11 
                 13 
                 13/13 = 1r0 
                 1 
               
               
                   
               
             
          
         
       
     
     The operation of the system corresponding to FIG. 7A is similar to that shown in FIG  6 B, However, the modulus circuit output now takes on 13 distinct values rather than only  5 . The number of values corresponds to the value of the denominator of the fractional value the system is attempting to achieve. Additionally, the PLL will now adjust the reference frequency two times every 13 cycles rather than two times every 5 cycles. 
     FIG.  6 A and FIG. 7A illustrate another feature of the present invention. As previously stated, FIG. 6A includes phase error tracking circuitry comprised of comparator  680  and exit state register  685 . As will now be explained, the phase error tracking circuitry of the embodiment illustrated in FIG. 6A will monitor the phase error of PLL  600  and reconfigure the PLL when the phase error is at a minimum. This allows the PLL system to acquire a signal in a high frequency fractional mode and then reconfigure the system into a lower frequency integer mode without loosing lock on the signal. FIG. 7B shows the actual phase error versus time of the PLL system for the fractional divisor ratio of N.F=1001 2/13. As FIG. 7B illustrates, the actual phase error of the PLL is highly non-linear. However, FIG. 7B illustrates that the phase error can be associated with corresponding results of the modulus circuit operation. As shown in FIG. 7B, each modulus result corresponds to an actual phase error. Therefore, it can be seen that the phase error of the PLL system of FIG. 6A, when configured to operate with a system divisor ratio N.F=1001 2/13, has minimal values during the phase detector cycles when the result of the modulus operation is either A modulus B=5 or A modulus B=4. If the system is reconfigured into a lower frequency integer mode during the phase detector cycle that has the least phase error, the PLL will be able to maintain lock after the reconfiguration. Therefore, a designer can simulate the system for a given set of parameters, determine which phase detector cycle has the minimum phase error, and use modulus results to control the reconfiguration of the system. 
     For the embodiment of FIG. 6A, a predetermined value is stored in the exit state register during system initialization. The PLL is then configured in a high frequency fractional mode. In a high frequency fractional mode the PLL will exhibit a faster lock time. After the PLL achieves lock, the comparator  680  monitors the result of the modulus operation and compares it to the predetermined value stored in the exit state register. When the PLL cycles through a point where the result of the modulus operation corresponds to a minimum phase error, the comparator signals the system to reconfigure into a lower frequency integer mode. 
     FIG. 8 illustrates one embodiment of a PLL system in a lower frequency integer mode according to the present invention. PLL  800  of FIG. 8 includes a phase detector  820 , a loop filter  830 , a VCO  840 , an R-divider  810  having a division ratio of R, and a reference denominator divider  811  having a division ratio of D. The output of the VCO Fout is divided down in the feedback path by a feedback denominator divider  812  having a division ratio of D, a B-divider  850  having a division ratio of B, an auxiliary divider  851  having a division ratio of A, a cycle slip controller  852 , and a prescaler divider  865 . PLL  800  also includes a numerator register  880  and a denominator register  870  for storing the fraction that corresponds to the fractional portion of the divisor ratio of the PLL when configured in fractional mode. 
     To understand the operation of the PLL when configured in integer mode, it is important to note first that the input reference frequency Fref and the PLL output frequency Fout are the same in both the higher frequency fractional mode and the lower frequency integer mode. Secondly, it is important to note that the input to the phase detector has been reduced by a factor of D, the value of the denominator of the fractional portion of the divisor ratio for the fractional mode. Because the frequency at the input to the phase detector has been reduced by a factor equal to the denominator value of the fractional portion of the divisor ratio of the PLL when operating in fractional mode, the PLL can maintain the same frequency at the VCO output with lower frequency at the input of the phase detector. This is illustrated by noting that if an extra K+1 cycle is introduced for every N out of D output pulses of the B-divider, then the total number of VCO output pulses required to generate one pulse at the phase detector input will be: 
     
       
         Ntot=N(BK+A+1)+D−N)(BK+A) 
       
     
     
       
         Ntot=DBK+DA+N 
       
     
     In other words, for one output pulse of the feedback denominator divider  812 , there will be a total of D pulses (or cycles) of the B-divider  850 . Additionally, for each cycle of the B-divider there will be (BK+A) cycles of the VCO output when the slip controller does not add an additional K+1 divide, and there will be (BK+A+1) cycles of the VCO output when slip controller adds an additional K+1 divide. Therefore, if an extra K+1 prescaler cycle is inserted in N (the numerator value) of the B-divider pulse cycles, then the number of VCO output pulses required to generate these N pulses is N(BK+A+1). Additionally, the number of VCO output pulses required to generate the remaining D−N) B-divider pulse cycles is D−N)(BK+A). As shown in the previous equation for Ntot, adding these together gives us the total number of VCO output pulses required to generate one pulse at the output of the feedback denominator divider  812 . Therefore, the VCO output frequency will be: 
     
       
         Fout=F2*(DBK+DA+N) 
       
     
     
       
         F1=Fref/R*D 
       
     
     
       
         F1=F2 assuming the PLL is locked 
       
     
     
       
         Fout=Fref (DBK+DA+N)/(R*D) 
       
     
     
       
         Fout=Fref D(BK+A)/R*D+(Fref/R) (N/D) 
       
     
     
       
         Fout=(Fref/R)(BK+A)+(Fref/R)(N/D) 
       
     
     This is the same output frequency as in fractional mode. However, the frequency at the phase detector has been reduced by a factor of D. 
     
       
         F 2_int=( 1/D)*Fout/((BK+A)+N/D) Integer Mode 
       
     
     
       
         F 2_frac=Fout/((BK+A)+N/D ) Fractional Mode   
       
     
     The lack of phase stepping in the input to the phase detector reduces the spurious tone power of the PLL. 
     Another embodiment of the present invention is shown in FIG.  9 . The PLL  900  of FIG. 9 is one possible implementation of the present invention that uses down counters for the dividers. PLL  900  receives a reference frequency Fref. Fref is applied to the strobe input of a down counter R-counter  915 , causing R-counter to decrement its value for every pulse of Fref. The output of R-counter  915  is transmitted to the strobe input of another counter Denom-Ctr  912 . Denom-Ctr  912  is programmed with the denominator value of the fractional system divisor ratio when the PLL is operating in integer mode as will be described in more detail below. The outputs of both the Denom-Ctr  912  and the R-counter  915  are received by two inputs of a 2-1 Multiplexor (“MUX”)  910 . The MUX  910  is controlled by a configuration signal received on its select input. When the PLL is in Fractional Mode, the configuration signal programs the MUX to transmit the output of the R-counter. When the PLL is in Integer Mode, the configuration signal programs the MUX  910  to transmit the output of Denom-Ctr  912 . The selected signal from the MUX  910  is transmitted to a synchronization circuit (“SYNC”)  908  having a input coupled to the output of the MUX  910 . The output of the SYNC  908  is transmitted to one input of phase detector  907 . PLL  900  also includes a charge pump  905 , a loop filter  903 , and a VCO  901  connected substantially as shown in FIG.  9 . The output of the VCO, Fout, is fed back to prescaler  940 . Prescaler  940  will divide the frequency of Fout by K or K+1 depending on the state of its control input, CTRL_IN. A logic low level at the CTRL_IN input of prescaler  940  will program the prescaler to divide by K, and a logic high level at CTRL_IN will program the prescaler to divide by K+1. The output of the prescaler  940  is received by the strobe input of another down-counter B-counter  914 . The output of B-counter  914  is received by yet another down-counter Denom-Ctr  913 . Denom-Ctr  913  is programmed with the denominator value of the fractional system divisor ratio when the PLL is operating in integer mode as will be described in more detail below. The outputs of both the Denom-Ctr  913  and the B-counter  914  are received by two inputs of a 2-1 Multiplexor (“MUX”)  911 . The MUX  911  is also controlled by a configuration signal received on the select input. When the PLL is in Fractional Mode, the configuration signal programs MUX  911  to transmit the output of the B-counter. When the PLL is in Integer Mode, the configuration signal programs the MUX  911  to transmit the output of Denom-Ctr  913 . The selected signal from MUX  911  is transmitted to a synchronization circuit (“SYNC”)  909  having a input coupled to the output of the MUX  911 . The output of SYNC  909  is transmitted to the second input of phase detector  907 . SYNC circuits  908  and  909  are strobed by Fref and the prescaler  940  output, respectively, in order to ensure synchronous transfer of pulses to the input of the phase detector  907 . 
     PLL  900  is initially configured into a Fractional Mode having a high frequency of pulses at the input of phase detector  907 . Such a configuration will allow the PLL to achieve a faster lock time. Fractional operation is achieved as follows. The output of prescaler  940  is transmitted to the strobe input of A-counter  916 . A-counter  916  is a down counter that will decrement every time it receives a pulse from prescaler  940 . When A-counter  916  reaches a count of zero, the signal at the CNT_ 0  output of A-counter  916  transitions to a logic high. This signal will be inverted in an inverter  944  and program prescaler  940  to stop dividing by K+1 and begin dividing by K. 
     In Fractional Mode however, some phase detector cycles must include an additional K+1 division in the prescaler cycle (as described with respect to PLL  600  of FIG.  6 ). To accomplish this the fractional portion of the system divisor, 0.F, is represented as a fraction N/D. The denominator D is programmed into the denominator register  917  and the numerator N is programmed into the numerator register  918 . During each phase detector cycle, the B-counter  914  output (which is also the input to the phase detector in Fractional Mode) triggers the accumulator  950  to add the value in the numerator register  918  to the result of a modulus circuit  955  output. The output of accumulator  950  undergoes a modulus operation with the value in the denominator register  917 . If the modulus circuit  955  overflows in a given phase detector cycle, the PLL is signaled to divide Fout by K+1 for one additional cycle of the prescaler  940  in the given phase detector cycle. This is achieved by programming the slip circuit  920 . Slip circuit  920  includes a multiplexor input  921 . The multiplexor  921  has one input connected to the output of the modulus circuit  955  and another input connected to a numerator counter  919 . When the system is configured in Fractional Mode, the select input of multiplexor  921  programs the modulus circuit OVF output to be transmitted to the slip circuit  920  input. On the other hand, when the system is configured in Integer Mode, the select input of the multiplexor circuit  921  programs the output of the numerator counter  919  to be transmitted to the slip circuit  920  input. When the modulus circuit  955  overflows in a given Fractional Mode phase detector cycle, the OVF output transmits a signal to the input of slip circuit  920 . When A-counter  916  reaches zero later in the phase detector cycle, the CNT_ 0  output will attempt to reprogram the prescaler  940  to divide by K. However, the CNT_ 0  will also trigger the OVF signal into the slip circuit  920  by strobing the CK input of the slip circuit  920 . This will activate the output of the slip circuit  920  and drive through OR gate  942  to program prescaler  940  to divide Fout by K+1 for one additional prescaler cycle. On the next prescaler cycle, the prescaler output pulse will reset the slip circuit  920 , and the prescaler  940  will be programmed to divide the output Fout by K, rather than K+1. In this manner, one additional division by K+1 is introduced into the appropriate phase detector cycles. This will result in a fractional relation between the frequency at the input of the phase detector  907  and the output Fout. 
     Once the system has achieved lock, comparator  960  is enabled (or clocked) by the phase lock indicator signal from the PLL lock detection circuit  1000  and compares each modulus result to the value programmed into the exit state register  975 . (It should be readily understood that, as one alternative to the PLL lock detection circuit  1000 , a counter can be used to count pulses, e.g., of the output signal from the VCO  901  or prescaler  940 , and after some predetermined number of such pulses the PLL is assumed to be locked. As another alternative, a counter can be used to count charge pump events, e.g., signal pulses at the output of the charge pump  905 . After the frequency of such events, independent of the frequency of operation of the PLL, has declined to some predetermined value the PLL is assumed to be locked.) Each modulus result corresponds to a certain phase error between the signals at the input of the phase detector. The value programmed into the exit state register will be the value which corresponds to the phase detector cycle with minimum phase error. This can be determined by the user by simulating the system for the fractional value desired. Techniques for simulating PLL systems and obtaining the corresponding phase error characteristics are well known to those skilled in the art. When the modulus circuit outputs a modulus result equal to the value in the exit state register, the configuration output of the comparator  960  will be activated and cause the system to reconfigure into an integer mode of operation. In the PLL  900  of FIG. 9, reconfiguration is achieved by transmitting a configuration signal on the CFG output of comparator  960 . The configuration signal is transmitted to MUX  910 , MUX  911 , and MUX  921 , causing each MUX output to correspond to the B inputs rather than the A inputs. This system will now operate in Integer Mode as will now be described. 
     When the configuration signal is transmitted to MUX&#39;s  910  and  911  the input of the phase detector  907  receives the output of Denom-Ctr&#39;s  912  and  913 . Both of these counters  912 ,  913  are loaded with a value from the denominator register  917 . 
     Therefore, because the B-counter output and R-counter output are now the Denom-Ctr inputs, both inputs of the phase detector  907  are effectively reduced by a factor of D, the value in the denominator register  917 . Additionally, the VCO output frequency is maintained by adding an additional K+1 prescaler division cycle in N out of D cycles of the B-counter  914  as previously discussed with respect to FIG.  8 . The additional K+1 prescaler division cycles are introduced into N out of D B-counter cycles by loading the numerator counter  919  with the value of the numerator register  918  at the beginning of a phase detector cycle. When signaled by the CFG output of the comparator  960 , MUX  921  will select the numerator register output CNT_ 0 * as the input to the slip circuit. CNT_ 0 * is logic high when the count is not equal to zero. 
     When the system is in lock, each phase detector cycle will include a number of B-counter cycles. The B-counter cycles will be determined by the output of the prescaler  940 . For the first N B-counter cycles, the value in the numerator counter  919  will be non-zero and CNT_ 0 * will be at a logic high. This value will be loaded into the slip circuit  920  at the end of every A-counter cycle. The result will be one additional K+1 division in the prescaler  940  for each B-counter cycle until the numerator counter  919  has counted down to zero. The count will reach zero and CNT_ 0 * will go low after N cycles of the B-counter  914  (where N is the value programmed in the numerator register  918  and therefore the value in the numerator counter  919 ). For the remaining D cycles of the B-counter  914  that are required to produce a pulse at the output of the Denom-ctr  913  (where D is the denominator value), CNT_ 0 * will be logic low, and an additional K+1 division will not be introduced into the B-counter cycle. As shown by the equations above for FIG. 8, the PLL output frequency Fout, which is also the output of the VCO  901 , will be the same as in Fractional Mode. 
     PLL  900  of FIG. 9 also illustrates another feature of the present invention. 
     PLL  900  further includes a CP register  970 . CP register  970  is programmed with an optional charge pump correction flag. When the output of the modulus circuit  955  produces a value corresponding to the minimum phase error, the PLL according to the present invention will reconfigure itself from Fractional Mode into Integer Mode. However, as shown in FIG. 7B, the actual phase error corresponding to a minimum modulus circuit output is not exactly zero. Therefore, CP register  970  is included to signal comparator  960  to output a charge pump correction signal on the CP output. The charge pump correction signal is used to adjust the performance of the charge pump  905  during a transition from Fractional Mode to Integer Mode. 
     In one embodiment, the charge pump correction signal programs the charge pump  905  directly to reduce the gain of the charge pump  905  by one-half when the system is reconfigured from Fractional Mode to Integer Mode. As illustrated by FIG. 7B, a normal charge pump pulse would cause a change in the output of the VCO  901  that would result in a phase error change represented by A. By reducing the gain of the charge pump  905  by one-half, the output shift will be reduced. Therefore, when the PLL enters Integer Mode, the initial phase error at the input of the phase detector  907  will be reduced. 
     Another embodiment of the charge pump correction technique is illustrated in FIG.  9 . PLL  900  of FIG. 9 also includes a digital-to-analog converter (“DAC”)  930  and a frequency correction register  935 . Prior to programming the system, the user of the present embodiment will simulate the phase error as previously described and determine the amount of phase error that will be at the input of the phase detector  907  during a transition from Fractional Mode to Integer Mode. The frequency correction register  935  is then programmed with a value corresponding to a DAC  930  output that will adjust the characteristics of the charge pump  905  output such that the VCO  901  will shift to a position of minimum phase error in Integer Mode. In other words, when PLL  900  is operating in Fractional Mode the charge pump  905  is producing an output which adjusts the output of the VCO  901  such that the phase error at the input of the phase detector  907  follows a pattern as shown in FIG.  7 B. When the system transitions into Integer Mode, the phase error correction signal is received on the enable, EN, input of the DAC  930 , and the DAC  930  outputs a signal of predetermined magnitude to adjust the output of the charge pump  905  such that the resulting shift in the VCO  901  output yields a minimum phase error at the input of the phase detector  907  when the system is in Integer Mode. The output signal of the DAC  930  can be a current output or a voltage output depending on the specific implementation of the charge pump  905 . Many charge pump designs are known in the prior art, and it would be evident for a skilled person in the art to determine, based on the architecture of the charge pump, whether to use a current output or voltage output DAC as well as the details of how to implement the adjustment. 
     For example, the enable signal EN of the DAC  930  can be set to cause the system to operate in open loop mode, e.g., whereby the appropriate voltage is applied to the loop filter  903  to ensure minimum phase error and the loop is “opened” by causing the output terminal of the phase detection stage  906 , i.e., the charge pump output, to enter a high impedance state of operation. Meanwhile, leakage of charge from such output terminal normally caused by the internal output stage is prevented by sampling the voltage at the output terminal immediately following the point in time when the output terminal is placed into its high impedance state. This sampled voltage is stored across a capacitive circuit element, buffered by a buffer amplifier and fed back to the output terminal during the holding period. A more detailed description of this circuit operation can be found in commonly assigned, copending U.S. patent application Ser. No. 09/383,162, filed Aug. 25, 1999, and entitled “Voltage Sample and Hold Circuit for Low Leakage Charge Pump” (the disclosure of which is incorporated herein by reference). (Open loop modulation techniques are used in a number of wireless communications systems, such as DECT, Bluetooth (http://www.bluetooth.com) and HomeRF (http://www.homerf.org) systems.) 
     In the foregoing specification, the invention has been described with reference to specific exemplary embodiments thereof. It will, however, be evident that various modifications and changes can be made thereto without departing from the broader spirit and scope of the invention as set forth in the appended claims. The specification and drawings are, accordingly, to be regarded in an illustrative rather than restrictive sense.