Abstract:
A system for locating an impairment in a coaxial cable network comprises an encoder, an impairment detector, and a decoder. The encoder couples to the network at a predetermined encoding point, upstream of the impairment. The encoder automatically encodes an identification code on a signal originating downstream of the encoding point and associated with the impairment. The impairment detector couples to the network at an access point, upstream from the encoding point, and receives signals from the network. The detector is adapted to detect from the received signals the signal associated with the impairment and generate a detected version of the signal. The decoder is adapted to decode the identification code from the detected version of the impairment signal. Once the identification code is determined, the encoder and encoding point are identified, and the location of the impairment is determine to be downstream of the encoding point.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application claims the benefit of U.S. Provisional Application No. 60/945,094, filed Jun. 19, 2007. 
    
    
     BACKGROUND OF THE INVENTION 
     The present invention relates generally to network monitoring systems, and more particularly to methods and apparatus for locating impairments in a hybrid fiber-coax (HFC) network. 
     Two categories of impairments regularly present themselves within HFC networks. They are (1) common path distortion (CPD), and (2) noise. Sources of noise may be internal to the network or external to the network. The latter is called “ingress”. Transmission difficulties of voice and data traffic over the return path band in an HFC network, due to CPD and/or noise, are well known and have been well documented in the past ten to fifteen years. Over time, in order to meet the increasing demands of more traffic on their networks, operators have moved to higher throughput formats for return transmission from home, e.g., 16 QAM. These higher order modulation signals are inherently more sensitive to CPD as well as noise. Error correction and other schemes have been implemented to minimize this sensitivity, however the problem still exists. Excessive CPD and noise in the return path can cause disruption to services such as voice and data. As such, it is important and of value to the operator of such networks to be able to quickly locate sources of these impairments such that the problem can be repaired and customer disruption can be minimized. 
     There are several strategies employed by operators in an attempt to manage return path impairments on the network as well as methods used to try to find and fix problems when they occur. 
     With the goal of managing ingress, some operators choose to install devices such as passive filters (high pass filters, window filters, or step attenuator filters) that reduce ingress coming from the subscriber&#39;s home. Another approach is to install Dynamic Ingress Blocking (DIB) systems that attenuate all or portion of the return path signals during periods of idle traffic and allow all signals through when traffic is present. This latter approach effectively blocks ingress when no traffic is present and allows it to enter the network when traffic is present. It must be noted that these approaches are not completely effective and they do not attempt to locate or eliminate sources of noise and ingress. Many operators alternatively choose not to use any noise mitigating devices. 
     In order to find and fix problems, many operators have installed return path spectrum analysis monitoring systems. When a noise or ingress problem is measured at the headend, a typical process is for a technician to go into the field and troubleshoot directly by temporarily disconnecting a portion of the return plant while simultaneously monitoring the effect this has at the headend. If ingress or other noise decreases at the headend when a return leg is pulled, a conclusion can be made that this was the direction in which the ingress or other noise was coming from. This trial and error process continues until the source of noise or ingress is found, which can be extremely time consuming. 
     Other employed systems include the Hunter® System by Arcom Digital, LLC, Syracuse, N.Y., described in U.S. Published Application No. US 2006/0248564 A1, published on Nov. 2, 2006, or one described in U.S. Published Application No. US 2004/0245995 A1, published on Dec. 9, 2004. These systems use passive detection techniques and correlation processing to compare simulated CPD signals generated from forward path signals at the headend to signals measured from the return path. These systems can pinpoint sources of CPD in an HFC network by calculating the distance to the source from a time delay determined from the correlation processing. However, due to inherent inaccuracies of network maps utilized in the cable television industry, further efforts to locate the source are often required. These additional efforts can be time consuming. 
     When an impairment is measured at the headend, the network operator can use a variety of methods in an attempt to find the cause of the problem. In a typical node on an HFC network, there are many branches and hundreds of devices, each of which could be the cause of the problem. Localizing the source of the problem can be a time consuming endeavor. Some cable systems utilize low attenuation value switches (termed “wink” switches) that attenuate ingress and other noise signals in various portions of the plant to assist the user in localizing the source of noise and ingress. Each wink switch has a unique address, and the various switches within the node are sequentially addressed and turned on such that a few dB of additional attenuation is introduced onto the return path of the leg of the network where the switch is attached. The return path spectrum at the headend is monitored while this switching is occurring—thereby pointing to a particular leg if the timing of when a switch is turned on corresponds to a noise level drop by a corresponding amount at the headend. Another technique employs a field spectrum analyzer in an attempt to troubleshoot the source. Other system operators use dipole antennas installed on trucks that are driven around the system in an attempt to triangulate a source. Others may use the previously mentioned Hunter® System. 
     The methods described above have deficiencies. In either the DIB or wink switch approach, it is necessary to introduce additional carriers transporting data over the return path in order to address the devices. In addition, the addressable devices installed in the field are complicated and relatively expensive. In the case of the DIB approach, it is required that additional devices be installed in the cable network. This is an expensive procedure that requires system downtime and service interruption. These additional devices also become potential new sources of network problems. The power consumption of such devices can make them unusable in some applications, or could necessitate relocation or adding power supplies to the network. In the case of wink switches, they are only installed in certain active devices; thus, the number of such switches may be limited in the network. Finally, in the case of wink switches, the amount of attenuation of the switch (typically 3 or 6 dB) could be disruptive to network traffic. 
     In this specification, the term “encoder,” “encoder device,” “ID encoder” or “probe” refers to a device that is installed in the cable network at an appropriate location, and which imparts or encodes an ID code, e.g., a frequency division or code division ID (as further described below), on the signal traffic in the network, so that the location of the device and of any impairments originating downstream from the device can be determined. The device is also sometimes referred to as a “marker” or “marker device” because it can be installed to mark or flag the network leg or branch from which the impairment originates. 
     OBJECTS AND SUMMARY OF THE INVENTION 
     It is therefore an object of the present invention to provide methods and apparatus for locating impairments in HFC cable networks that overcomes the problems associated with the prior art. 
     It is another object of the present invention to provide methods and apparatus for locating network impairments more accurately and reliably than previous methods and apparatus. 
     It is a further object of the present invention to provide methods and apparatus for locating network impairments that do not require the transmission of additional signals in the return path frequency band. 
     It is a still another object of the present invention to provide methods and apparatus for locating network impairments that do not require addressable devices deployed in the network. 
     It is still a further object of the present invention to provide methods and apparatus for locating network impairments that include encoder or marker devices deployed in the network that are easy to install, and the installation does not require the disconnection of the network or disruption in service to subscribers. 
     It is yet another object of the present invention to provide methods and apparatus for locating network impairments that include encoder or marker devices that are of relatively simple design and low cost. 
     It is yet a further object of the present invention to provide methods and apparatus for locating network impairments that include encoder or marker devices, the operation of which do not disrupt or interfere with service in either the forward or return paths of the network. 
     These and other objects are attained in accordance with the present invention wherein there is provided a system of locating impairments in an HFC network, comprising: (a) a network impairment detector that determines whether an impairment, such as ingress or CPD, is present in the network; (b) at least one encoder or marker device deployed in the network at a known encoding point, which encodes return path signals (including signals generated by impairments) originating downstream of the encoding point with an ID code; and (c) a decoder unit that decodes the ID code and identifies the encoder and the encoding point (i.e., the encoder&#39;s location), whereby it is determined that any network impairments detected by the impairment detector are downstream of the encoding point. 
     In the preferred embodiment, the encoder or marker device is in the form of a threaded probe that screws into an access port (or seizure port) of a splitter, multi-tap, amplifier or other passive or active network device already installed in the network. The probe makes contact with a seizure screw in the device. Thus, the preferred method of installation does not require any cutting of cables or disruption of service. The preferred method of encoding is amplitude modulation (AM) of the return signals with a single frequency, periodic function. In this case, the decoder unit is an amplitude demodulator. In the preferred embodiment, the decoder unit is combined with the impairment detection unit. 
     Methods of locating impairments in an HFC network are also contemplated by the present invention. One such method comprises the steps of: (a) placing an ID encoder at an encoding point in the network, wherein the network has forward and return path frequency bands; (b) placing a network impairment detector and an ID decoder at a test point in the network, upstream from the encoding point; (c) imparting an ID code from the encoder on signals in at least a portion of the return path frequency band, which signals originate downstream from the encoding point; (d) receiving the encoded signals at the test point; (e) determining whether an impairment is detected from the encoded signals, in the network impairment detector; and (f) decoding the ID code with the ID decoder to identify the ID encoder and the encoding point in the network, whereby any impairment detected in the previous step is determined to be downstream of the encoding point. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWING 
       Further objects of the present invention will become apparent from the following description of the preferred embodiment with reference to the accompanying drawing, in which: 
         FIG. 1  is a block diagram of a system of the present invention for locating impairments in an HFC CATV network; 
         FIG. 2  is a block diagram of a portable system of the present invention for locating impairments in an HFC CATV network; 
         FIG. 3  is flow diagram illustrating the preferred steps of carrying out the invention; 
         FIG. 4  is a perspective view of an encoder probe of the present invention; 
         FIG. 5  is a perspective view of the encoder probe threaded onto an access port of a line splitter, to illustrate the ease of installation of the probe into the coaxial portion of the network plant; 
         FIG. 6  is a lateral cross sectional view of the probe, illustrating its construction;  FIG. 6A  is cross sectional view taken along line A-A′ in  FIG. 6 ; and  FIG. 6B  is an enlarged cross sectional view of the probe; 
         FIG. 7  is a schematic diagram of an RF circuit installed in the probe of  FIG. 6 ; 
         FIG. 8  is a schematic diagram of a digital circuit installed in the probe of  FIG. 6 ; 
         FIG. 9  is a frequency response plot of the probe in  FIG. 6 ; 
         FIG. 10  is a schematic diagram of a coaxial portion of a network plant; 
         FIG. 11  is a block diagram of the signal processing steps for noise impairment detection performed in the systems of  FIGS. 1 and 2 ; 
         FIG. 12  is a block diagram of the signal processing steps for CPD impairment detection performed in the systems of  FIGS. 1 and 2 ; 
         FIG. 13  is a block diagram of a correlator used in CPD impairment detection in the systems of  FIGS. 1 and 2 ; and 
         FIG. 14  is a plot of the correlation function from the correlator of  FIG. 13 . 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
       FIG. 1  shows a block diagram of one embodiment of an impairment location system A of the present invention. System A is incorporated into a CATV headend B and a coaxial cable portion  10  of an HFC cable TV network. System A includes the following equipment connected at the headend: a headend impairment detection and decoding unit  1 , a return path switch  3 , and a computer  2 . System A also includes at least one ID encoder  12  connected to or integrated in coaxial cable portion  10  of the HFC network. The general headend equipment includes a headend combiner  4  for combining forward path signals (i.e., analog and digital TV program signals), a signal splitter  5  for delivering the forward path signals to a plurality of optical nodes, a plurality of optical transmitters  6  for transmitting the forward path signals to fiber optic nodes in the fiber optic portion of the HFC network, and a plurality of optical receivers  7  for receiving return path signals from the optical nodes and delivering them to headend receivers (not shown). 
     Headend impairment detection and decoding unit  1  has two inputs: one is connected to an output of return path switch  3 , and the other is connected to an output of combiner  4 , through a tap  11 . An output (“RF OUTPUT”) of unit  1  is connected to an input of combiner  4 . The analog and digital TV program signals of the forward path are applied to the other inputs of combiner  4 . The combined output of combiner  4  is connected to an input of splitter  5 . The outputs of splitter  5  are connected to the inputs of optical transmitters  6 . The forward path signals are transmitted from transmitters  6  over optical cables to corresponding optical nodes  8 , as shown in  FIG. 1 . As previously indicated, the output of combiner  4  is also coupled through tap  11  to an input of unit  1  (i.e., the “Forward Path Input”). 
     Management of unit  1  and switch  3  is carried out with the help of computer  2 . As shown in  FIG. 1 , computer  2  is connected to unit  1  and switch  3 . Computer  2  is programmed to control switch  3  and cause switch  3  to connect the return path signals from each optical receiver  7  to another input of unit  1  (i.e., the “Return Path Input”), separately, in a cyclic or programmed (or selective) manner. In addition, computer  2  performs processing and storage of information received from unit  1 . 
     With further reference to  FIG. 1 , the output of optical nodes  8  are connected to a coaxial cable plant  10  of the network. Nodes  8  convert the forward path optical signals to RF signals for transmission down coaxial cable plant  10  and convert the RF return path signals to the optical spectrum for transmission to optical receivers  7 . Coaxial cable plant  10  is generally arranged in a tree-and-branch pattern, between nodes  8  and a multiplicity of subscribers&#39; homes  10   a . Only a portion of the tree-and-branch structure of plant  10  is shown in  FIG. 1 . However, what is shown is representative of the entire network plant. As shown there are distribution amplifiers  10   b , splitters  10   c , and multi-taps  10   d.    
     In accordance with the invention, ID encoders  12  are placed at encoding points in coaxial cable plant  10 . Ideally, encoders  12  should be placed in each terminal branch of plant  10 . However, this may not be possible due to the power requirements of encoder  12 , which will be explained further below. Encoder  12  is preferably in the form of a probe that is configured to be threaded into an access port of a system passive device, such as a splitter or multi-tap. The probe embodiment will be described in greater detail below. Encoder  12  can also be incorporated inside a system passive device as part of its circuitry. 
     As shown in  FIG. 2 , impairment detection and decoding unit  1  may be implemented as a portable or handheld unit  20 . This is the preferred form for the impairment detection and decoding unit of the present invention. As illustrated in  FIG. 2 , unit  20  can be connected to an RF test point in headend return path optical receiver  7  or at any test point (or other access point) in coaxial cable plant  10 , upstream of encoders  12 . 
     The general steps of the preferred method of carrying out the invention are shown in  FIG. 3 . In a first step  30 , at least one ID encoder  12  is placed or installed at an encoding point in coaxial plant  10 . The encoding point is preferably at the head of a branch in the plant, as shown in  FIG. 1 . A second step  32  includes placement or connection of impairment detection and decoding unit  20  at any test point at the headend or in coaxial plant  10 , as shown in  FIG. 2 . A third step  34  involves imparting an ID code from encoder  12  on signals (which includes noise) originating downstream of the encoding point, in at least a portion of the return path frequency band. The ID code is preferably in the form of amplitude modulation with the frequency of modulation being the identifying code. A fourth step  36  includes receiving the encoded signals at the test point. A fifth step  38  includes determining whether an impairment is detected from the encoded signals using unit  20 . The impairments to be detected may by common path distortion (CPD), noise and ingress. A sixth step  40  includes decoding the ID code using unit  20  to identify ID encoder  12  and its encoding point in coaxial plant  10 . The decoding portion of unit  20  is an amplitude demodulator if the ID code is implemented as amplitude modulation. The decoding function is preferably implemented in a digital signal processor (DSP) chip or in software on a special or general purpose computer (See  FIGS. 11 and 12 ). 
     As indicated before, ID encoders  12  are preferably configured in the form of a probe. In  FIG. 4 , a probe  42  constructed in accordance with the present invention is shown. Probe  42  comprises a generally cylindrical housing  44 , including a threaded connector  46 , and a probe tip  48 . Probe tip  48  includes a metal seizure contact  50  and an insulator assembly  52 . As shown in  FIG. 5 , probe  42  is threaded into an access port  54  of a splitter  56 . Splitter  56  is shown as an example only. Probe  42  can be connected to any type of passive or active device having an unused seizure or access port where contact to one of the device&#39;s seizure screws can be made. Splitter  56  includes cable ports  58   a - d .  FIG. 5  illustrates the simplicity of installation of probe  42  into network plant  10 . When probe  42  is fully threaded into access port  54 , metal contact  50  makes direct contact with the seizure screw inside splitter  56 . 
     The construction of probe  42  is shown in  FIGS. 6 and 6A . Housing  44  includes a body  60  and an end cap  62 , both made of nickel-plated brass or similar conductive metal typically used for CATV trap filter housings. Threaded connector  46  extends from body  60  and contains external threads  64  which are configured and dimensioned to thread into a seizure or access port on a network device. Insulator assembly  52  includes a generally cylindrical insulator body  66  and a cone-shaped insulator cap  68 , both made of polypropylene. Insulator body  66  includes a generally circular rear flange  70  and a cylindrical tail section  72 . Probe tip  48  is inserted into connector  46 . Flange  70  is compressed during insertion and acts as a stop against a shoulder  74  of probe body  60  (further described below). Four longitudinal grooves are contained on the outer surface of insulator body  66  to facilitate insertion into connector  46  and establish a firm but sliding fit. An o-ring  76  is seated in a circumferential groove between connector  46  and probe body  60 . O-ring  76  is an elastomeric seal and functions to seal the connection between probe  42  and the access port into which probe  42  is threaded ( FIG. 5 ). 
     A digital circuit board  78  and an RF circuit board  80  are mounted in housing  44 . The circuitry on boards  78  and  80  are not shown in  FIGS. 6 and 6B , but will be described below with reference to  FIGS. 7 and 8 . Boards  78  and  80  are disc-shaped. Boards  78  and  80  are connected together by means of a board interconnect  82  (see also  FIG. 6A ) and are spaced apart by a brass spacer  84 . Board  78  is soldered to end cap  62  and spacer  84  by means of a solder ring  86 . RF board  80  is soldered to end cap  62  and probe body  60  by means of another solder ring  88 . An LED  90  is mounted on the right side (as viewed in  FIG. 6B ) of digital board  78  and functions as an indicator light to indicate that probe  42  is in electrical contact with the seizure screw and is working properly. An aperture  92  is contained in a rear end wall  63  of cap  62  and is in line of sight with LED  90 , to allow light from LED  90  to be emitted through cap  62 . A transparent lens  94  seals aperture  92 . 
     The electrical contact assembly of probe  42  is best shown in  FIG. 6B . The contact assembly comprises seizure contact tip  50  made of nickel-plated brass, a leaded ferrite bead  96 , a dual-socket female conductor assembly  98 , and a male pin conductor  100 . Contact tip  50  is partially mounted in a cylindrical hole  67  through insulator body  66 . Ferrite bead  96  and conductor assembly  98  are mounted in hole  67 . Contact tip  50  is secured in place by insulator cap  68 , which engages a circumferential groove or shoulder in contact tip  50 . The left lead (as shown in  FIG. 6B ) of ferrite bead  96  is cut short and soldered inside contact tip  50 . Male pin conductor  100  is made from the same wire used for a male conductor or stinger of a CATV trap filter. Conductor assembly  98  includes two multi-point spring contacts  102  and  104 , press fitted into corresponding hollow (“female”) conductor pins  106  and  108 , respectively. Contacts  102  and  104  are the same as the contacts shown and described in U.S. Pat. No. 6,674,343. As seen in  FIG. 6B , contacts  102  and  104  have their insertion ends in opposite directions. The right lead ( FIG. 6B ) of ferrite bead  96  is inserted into and held by contact  102 . Female conductor pins  106  and  108  are coupled together by a cylindrical nickel-plated brass tube  110 . Pins  106  and  108  are preferably soldered to tube  110 . As shown in  FIG. 6B , conductor assembly  98  is generally hollow through and through. Male pin  100  has an output end  112  fixedly mounted to RF circuit board  80 , and an input end  114  inserted into contact  104 . Thus, an electrical connection is established between a seizure screw of a network device, into which probe  42  is threaded, and RF circuit board  80 . 
     Experience has taught that the seizure or access ports on various network devices have different lengths. Therefore, the length of probe tip  48  would need to be reduced or extended depending on the length of the ports. This would require the manufacture and inventorying of different length probes. Such a requirement would not be an ideal situation for the manufacturer or network operator. The probe of the present invention overcomes this problem by spring loading probe tip  48 , so that it can self-adjust to the correct length upon being threaded into the seizure or access port. As shown in  FIG. 6B , a spring  116  is mounted directly to tail section  72  at one end, and at its other end, spring  116  is seated in a cylindrical recess  117  contained in a moisture seal  118 . Spring  116  imposes an outward force on tip  48  (to the left in  FIG. 6B ). Insulator body  66  is in sliding engagement with the interior wall surface of connector  46 . Insulator body  66  is pushed outward by spring  116  to a point where it is stopped by engagement of flange  70  with shoulder  74 . When probe  42  is threaded into an access port that has a length shorter than the fully extended tip  48 , tip  48  is pushed inward (to the right in  FIG. 6B ) against the force of spring  116 . Male pin  100  remains stationary and conductor assembly  98  slides over male pin  100 , as tip  48  is pushed inward. The sliding engagement of conductor assembly  98  over male pin  100  is made possible because assembly  98  is generally hollow. 
     As shown in  FIG. 6B , moisture seal  118  is press fitted into an opening  120  contained in probe body  60 . Seal  118  and opening  120  each have two corresponding diameters. The larger diameters of seal  118  and opening  120  allow for a secure seating of spring  116  in recess  117 . The two diameters of seal  118  and opening  120  also aid in effecting a good moisture seal. Seal  118  is made of a low density polyethylene, or a polypropylene, or other appropriate seal material. Seal  118  seals the interface between male pin  100  and seal  118  and the interface between probe body  60  and seal  118 . 
     Probe  42  serves to mark the location in coaxial plant  10 , downstream of which the impairment is located. Probe  42  introduces a very slow and insignificant amount of attenuation (in the range of about 0.2 dB to about 2 dB) over a limited portion of the return path frequency band (e.g., 5-20 MHz). The slow modulation period and small attenuation value make its presence insignificant and not intrusive to any return path services. Preferably, probe  42  should only be installed in parallel branches of the network—this eliminates cascading of probes and the accumulation of return path attenuation by cascaded probes, which could possibly have a detrimental effect on the network. 
     Powering of probe  42  comes from the AC line power used to power coaxial plant  10 . Each probe  42  employed in a node of plant  10  has a unique characteristic or personal identification code. This personal identification code can be formed in a variety of ways. For example, if simple frequency division is use, each probe  42  could have its own assigned amplitude modulation (AM) frequency. As an alternative, code division or a combination of methods could be used. Frequency division methods are simple to realize, however the number of channels is limited (which therefore limits the number of personal identification code numbers). Further, the frequency division method demands lengthy signal recognition to achieve the required resolution. Code division techniques, such as the Gold or Kasami sequences could be used to create a larger number of ID combinations with minimal code length and minimum decoding correlation complexity. As an example, a 63 bit code length Kasami sequence will allow up to 520 unique ID codes. The choice of which code technique to adopt for the identification code depends upon practical considerations such as, e.g., cost, complexity, and the number of probes or encoders to be deployed in a network or node. 
     The schematic of the circuitry mounted on RF circuit board  80  is shown in  FIG. 7 . The signals received by probe  42  (through contact tip  50 ) enter the circuitry on board  80  through an input  124 . A shunt branch, comprising gas tube  126  and ferrite beads L 4  and L 5 , is connected between input  124  and ground. Gas tube  126  functions to protect the circuitry from power surges. Ferrite beads L 4  and L 5  are required to handle current surges and reduce glitches at high frequencies caused by gas tube  126 . A series inductor L 3  is connected between input  124  and a blocking capacitor C 6 . Inductor L 3  provides high frequency isolation to the circuit. An inductor L 2  and capacitor C 4  function as a lowpass circuit allowing AC power to pass to an isolation resistor R 2  and to an output  125 . The AC power signal from output  125  is routed through interconnect  82  to digital circuit board  78 . As shown in  FIG. 7 , a microprocessor controlled switch  127  is connected between blocking capacitor C 6  and a blocking capacitor C 3 . A capacitor C 5  is connected to the RF 1  pin of switch  127 . 
     An impedance circuit  128  is connected to blocking capacitor C 3 . Impedance circuit  128  includes a lowpass filter, C 1 , L 1  and C 2 , and a resistor R 1 . Impedance circuit  128  imparts a small attenuation (e.g., 0.5 dB) over a portion of the return path frequency band. The portion of the return path frequency band over which the attenuation is imparted is defined by the lowpass filter, C 1 , L 1  and C 2 . In one embodiment, the portion of the return path frequency band is about 5-20 MHz. Impedance circuit  128  effectively loads down (e.g., by 0.5 dB) the network downstream of the point where probe  42  is connected, in the selected portion of the return path frequency band (e.g., 5-20 MHz). Switch  127  is opened and closed under the control of a microprocessor located on digital board  78 , at a periodic rate uniquely assigned to the particular probe  42 . This rate is the ID code for the probe (i.e., the ID encoder). When switch  127  is closed, attenuation circuit  128  loads down the return path signals. When switch  127  is opened, the return path signals are unaffected. 
     A typical frequency response plot of probe  42  is shown in  FIG. 9 . The y-axis is attenuation in tenths of a dB, and the x-axis is frequency from 0-100 MHz. Curve  130  is the response of probe  42  when attenuation circuit  128  is switched into the network and curve  132  is the response of probe  42  when attenuation circuit  128  is switched out of the circuit. 
     The schematic of the circuitry mounted on digital circuit board  78  is shown in  FIG. 8 . The AC power signal from RF circuit board  80  is received through input  129 . A rectifier circuit  131 , comprising resistor R 9 , capacitor C 1 , diode U 1 , capacitor C 2 , resistor R 3 , and capacitor C 3 , converts the AC power signal to a rectified DC signal. A DC-to-DC converter  133  is connected between the output of rectifier circuit  131  and a DC filter circuit  135 , comprising capacitors C 4  and C 5 , resistor R 4 , and capacitor C 6 . A microprocessor  137  is connected at the output of DC filter circuit  135 . LED  90  is powered by microprocessor  137  through a bias resistor R 5 . The 60 cycle AC power signal is drawn through resistor R 2  and brought to a signal generating circuit  139 , comprising a zenor diode U 4 , a capacitor C 7 , and resistors R 6  and R 7 . Signal generating circuit  139  produces a positive sine wave signal that is received in pin  6  of microprocessor  137 . The sine wave signal is applied to the input of a Schmitt trigger circuit integrated inside microprocessor  137 . The Schmitt trigger circuit produces clock pulses for the microprocessor. A clock output signal is presented at pin  7  of microprocessor  137  and is routed to a clock port CLK of board interconnect  82  by means of output resistor R 8 . The control signals for opening and closing switch  127  (on RF board  80 ) are presented at pins  2  and  3  of microprocessor  137 . These control signals are transmitted to RF board  80  via connections TP 2  and TP 3  of board interconnect  82 . Microprocessor  137  is programmed to cause switch  127  to switch at the assigned ID rate. 
     Under some circumstances, it may be desirable to integrate the circuitry of probe  42  into a network device like a splitter, multi-tap, or amplifier—which would be a relatively easy undertaking. 
     When probe  42  is connected to a network device, it creates a condition wherein any signals, including CPD or noise signals, will be similarly modulated in the range of frequencies where the probe circuitry is switched. If there are no signals present, then the noise floor will be modulated. This modulation is so slow and of such low attenuation value that its effect on the transmitted signals (e.g., return path traffic) will be insignificant. It has been determined that the attenuation can be as low as 0.2 dB and achieve reliable decoding of the modulation frequency. 
     At the detection and decoding location (either at the headend or in the field), the modulation is detected and the individual identification code is resolved. Once the code is resolved it is a simple task to look up records of where the device with this code is currently installed, allowing the technician to greatly simplify the required troubleshooting process. This technique can be used to locate noise and ingress. Alternatively it can be used with other equipment like the Arcom Digital Hunter® System to resolve the location of CPD, which like the ingress will be similarly modulated. If either CPD or ingress is modulated in a fashion consistent with an installed probe (encoder), the source of the impairment generating the CPD or ingress must be located downstream of the probe (encoder). There can be multiple encoder devices installed in a node, each requiring a unique modulation frequency or other code ID. If it is desired that the measurement be done in the field, it is possible that duplicate codes could be used in a node if the duplicate devices are located in different legs of the plant and separate measurement points for the different legs are accessible. 
     The number of probes or encoders deployed can vary depending upon preferences of the operator. A reasonable number for the purposes of this system would be to place a probe at the first network device after the last active (i.e., amplifier) in each branch of the network. Because of additive losses, added complexity in resolving the probe IDs of cascaded probes, and uncertainty related to return loss of cascaded probes—it is not recommended that multiple probes be cascaded. While, technically, it creates no insurmountable problem to cascade probes, it could create situations where multiple probes in series could adversely affect the network performance. Since probe  42  requires network power, the placement of probes  42  may be predicated upon the location of the most downstream or last powered device within the network. As such, it may not be possible to place one probe in each leg. An analysis of typical node architectures in modern HFC plants shows that a typical number of probes placed in a node would be approximately 20-40. Typical points of installation are shown in  FIG. 10 . In  FIG. 10 , coaxial cable network plant  10  contains five probes  42 , installed in network branches  134 ,  136 ,  138 ,  140  and  142 , respectively. 
     When a probe is attached to a device, the return path noise floor and return path traffic downstream of the device will become modulated with a unique characteristic corresponding to the particular period of the modulation, that which we refer to as the device&#39;s personal ID. When ingress or CPD is present in the network downstream of the location of the probe, the ingress or CPD will be similarly modulated. Using signal processing techniques contained within equipment located upstream of the probe, the signal is demodulated or decoded such that the modulation frequency or code can be determined and then compared to the personal ID codes of all the probes contained in the node. The physical zone or part of the network from which the CPD or ingress is originating can then be easily found. Each installed probe will therefore be a marker, clearly pointing to where in the network ingress or CPD is originating. Each device will operate independently with no requirement of control or addressability. 
     Several methods of implementing the system can take place depending upon the desired level of automation. Systems could be employed that automatically monitor nodes and calculate personal ID codes for probes located in the plant upon CPD or ingress exceeding threshold levels. A more practical approach and the preferred embodiment is for a return path monitoring system to be employed at the headend. When a CPD or noise condition is seen on a node, a technician then uses resolving or decoder equipment either at the headend or at a location in the plant in order to resolve the individual ID code. 
     A block diagram of a device  150  used for ingress signal processing is shown in  FIG. 11 . Device  150  is first connected to the network through a return test point. The signal is received via an input  152  and then amplified by an amplifier  154 . The amplified signal is then filtered by low pass filter  156 . The filtered signal is then digitized by an analog-to-digital converter (ADC)  158  and then filtered by a digital bandpass filter (BPF)  160 . The passband of BPF  160  is less than or equal to the frequency band in which encoder  12  (or probe  42 ) modulates the ingress. Therefore if the AM modulation was implemented in the range of 5-20 MHz, BPF  160  could also be 5-20 MHz. If data or telemetry or other signals or carriers are used by the operator within this same frequency band, then BPF  160  could be made narrower to filter out these signals. The passband of BPF  160  is selected such that useful signals (from the cable operators&#39; perspective) are filtered out, and in the case of using the equipment to search for ingress—the maximum amount of ingress modulated from encoder  12  (or probe  42 ) remains. In the case of using the equipment to more narrowly search for CPD, BPF  160  is only required to pass the return frequency used by the Hunter® System to monitor for CPD. 
     When Resolving an Ingress Source: 
     After passing BPF  160 , the signals enter an integrator  162  which continuously sums the ingress samples squared, within certain time intervals. The signal at the output of integrator  162  looks like this: 
               S   ⁡     (   i   )       =       1   /   N     ⁢       ∑     j   =     k   *   i         N   +     K   *   i         ⁢       x   ⁡     (   j   )       2               
Where X(j) is the readout of a digital signal at the output of BPF  160  during the moments of time Tj, following the interval of digitization in ADC  158 , delta (Tj)=Tj−T(j−1); K is the integer factor defining the relationship between the signal digitization frequencies at an input and an output of integrator  162  (K&gt;1); and N is the quantity of signal X(j) samples accumulated in integrator  162 .
 
     Signal S(i) is a representation of the signal power, or more specifically the average energy of noise X(j) over a time interval of N samples. Thus integrator  162  represents a noise accumulator in which for digitized time intervals, the noise energy for a previous time interval N*delta (Tj) is continually estimated. If the given time interval is less than the period of the noise AM modulation from encoder  12  (or probe  42 ), then integrator  162  will work as an RMS detector. 
     Since signal S(i) is at a relatively low frequency, its sampling for the subsequent digital processing is done at a much lower frequency than the sampling of the direct noise source X(j). An example of the parameters of a practical realization of device  150  ( FIG. 11 ) is now given. For digitization of the 5-42 MHz return path signal of a typical HFC network in the United States, a 10-bit resolution and 100 MHz frequency can be chosen for ADC  158  (e.g., model AD9215 available from Analog Devices Company). BPF  160  and integrator  162  are easily realizable with a modern Field Programmable Gate Array (FPGA). Since integrator  162  works as a peak detector, the accumulation time in integrator  162  (or quantity of summed samples) should be chosen to be as long as possible, but less than the minimal period of AM modulation realized and implemented in encoder  12  (or probe  42 ). As an example, assume the minimal AM modulation period in encoder  12  (or probe  42 ) is 2 seconds. The accumulation time in integrator  162  can be chosen to be 0.5 seconds. Integrator  162  should therefore collect N=100^6*0.5 or 50^6 (fifty million) samples of noise signal X(j). 
     The output of integrator  162  represents a noise envelope AM modulated with a frequency of a single Hz. To digitize this low frequency signal it is sufficient to use a frequency of 20 Hz. Therefore, samples of the integrator output signal S(i) will be taken with a period of 50 milliseconds, while samples of noise enter integrator  162  with a period of 10 nanoseconds (100 MHz). This corresponds to an integer factor of K=5^6. Technical realization of these parameters is easily achievable. 
     The output signal of integrator  162  then enters a computer or a digital signal processor (DSP)  164 . The integrator output signal is of a rather low frequency; therefore the data transmission process will cause no problems. Computer  164  accumulates the data file sufficient for AM demodulation and the corresponding encoder  12  (or probe  42 ) ID code detection (i.e., decoding). 
     The minimal duration of the signal entering computer  164  for processing should be selected based primarily on ensuring reliable AM signal detection and selection of adjacent ID codes. For example, in the case of using frequency coding ID (code selection by AM frequencies) and a choice of 20 coding frequencies distributed over regular intervals, in a range from 0.5 up to 1 Hz, the frequency interval between adjacent codes will be 0.5 Hz/20=0.025 Hz. For division of such codes with the use of classical Rayleigh criterion, it is necessary to provide the minimal duration of signal analysis equal to 1/0.025=40 seconds. An increase in the number of codes will result in a proportionally increased time duration required for analysis of the received signal. Under real conditions within a cable television network, the AM signal is received on a background of various sorts of jamming signals caused by ingress signal instability; therefore, it will be necessary to increase the time duration of the received signal to ensure proper reception. From the point of view of a practical realization, it is reasonable not to fix rigidly the signal analysis time but only to limit it to some reasonable value, for example, several minutes. If within the limits of this time interval the AM code is detected by the chosen criterion (for example, signal-to-noise-ratio (SNR)), then the process of signal reception can be easily stopped. 
     When Resolving a CPD Source: 
     A block diagram of a device  170  for locating CPD sources is shown in  FIG. 12 .  FIG. 12  is very similar to the block diagram of the Arcom Digital commercial Xcor® radar and contains all of the same basic elements. See U.S. Published Application No. US 2006/0248564 A1 (published Nov. 2, 2006). The operational difference of using it for encoder  12  (or probe  42 ) detection (i.e., decoding) consists of adding an additional data processing algorithm implemented in a computer (or DSP). The device works as follows. Forward path signals enter device  170  at an input  172  and then are filtered by a bandpass filter (BPF)  174 , wherein a part of the forward path spectrum containing only QAM signals is selected. The selected QAM signals are amplified in an amplifier  176  and then enter a CPD emulator  178 . CPD emulator  178  represents a nonlinear element, for example, a diode from which second order products from the QAM signals are formed. Second order products from the QAM signals represent a noise signal with practically uniform spectral density in the range of return path frequencies (“formed noise signal”). By means of a bandpass filter BPF  180 , selection of a part of the formed noise signal spectrum is carried out. The part selected is a zone of frequencies where there are no return path service signals. Precisely the same part of the signal spectrum is allocated in the channel of return path signal reception (left-side channel in  FIG. 12 ). The signal at the output of BPF  180  will now be referred to as the “reference signal.” The reference signal is digitized by an analog-to-digital converter  182 . 
     As shown in  FIG. 12 , return path signals enter device  170  at an input  184  and are filtered by a bandpass filter (BPF)  186 . BPF  186  has the same frequency response as BPF  180 . The bandpass filtered signals are then amplified by an amplifier  188 . The signals at the output of amplifier  188  are filtered by lowpass filter  190  and then digitized by analog-to-digital converter  192 . These signals contain what we will term an “echo” signal from CPD sources (but in reality is not an echo, but second order intermodulation products from the forward QAM signals). The echo signal is the same as the reference signal, however, in the return path channel, the echo signal is delayed by time, corresponding to the time delay distance to the CPD source in the cable network. For echo signal detection in the return path, a digital correlation receiver  194  is used. The digitized versions of the reference signal and echo signal enter digital correlator  194  ( FIG. 12 ). 
     A block diagram of digital correlator  194  is shown on  FIG. 13 . Correlator  194  can be implemented in a field programmable gate array (FPGA). Correlator  194  has an input  197  (for receiving the reference signal) and N channels  198  with a uniform incremental delay To. Correlator  194  has an additional channel  199  with no delay. The reference signal is delayed by a minimum time delay in a variable delay line  200 . The minimum time delay is controlled by a control signal from a computer  196  ( FIG. 12 ). The control signal enters variable delay line  200  at an input  201 . Channels  198  and  199  each include a multiplier  202  and an integrator  203 . The echo signal from ADC  192  ( FIG. 12 ) enters correlator  194  at an input  204 . The echo signal enters an input of each multiplier  202  of channels  198  and  199 . The delayed reference signal from delay line  200  enters channels  198  and  199 . The delayed reference signal then enters multiplier  202  in channel  199  without a further delay, and enters multipliers  202  in channels  198  after being delayed by incremental delay lines  206 , respectively. Delay lines  206  introduce the incremental delay, N*To, as indicated above. The outputs of multipliers  202  are integrated in integrators  203 , respectively, and the results are stored in a register  208 . Control of the integration time of integrators  203  and the readout of integration results from register  208  are accomplished by a timing signal from computer  196 . The timing signal enters integrators  203  and register  208  via a control line  207 . The integration results from register  208  are directed to computer  196  via an output  209 . 
     The delay increment To of channels  198 , as a rule, corresponds to the period of digitization of signals in ADCs  182  and  192  ( FIG. 12 ). For example, choosing a frequency band for echo signal reception (i.e., passband of BPF  186 ) within the limits of 8-16 MHz, it is expedient to choose the frequency of digitization in ADCs  182  and  192  to be on the order of 40 MHz. Thus time To=25 nsec. The number of channels  198  in correlator  194  and the time delay in variable delay line  200  are selected so that a full possible range of time delays of echo signals from CPD sources is covered. For example, if the time delay in the optical cable portion of the HFC network (CPD cannot occur within fiber) is 100 micro seconds, and the maximum time delay within the coaxial portion of a particular node of the HFC network does not exceed 25 micro seconds, then for the guaranteed overlapping of possible times of arrival of the echo signals, it is expedient to choose a time delay in variable delay line  200  equal to 95 micro seconds, and the number of correlation channels equal to (10+25) Micro sec/To=(10+25)/0.025 =1400. 
     Voltage readouts from the outputs of each correlation channel are synchronized to the integration time in integrators  203 , and the readouts are then transferred to computer  196  ( FIG. 12 ). It has been experimentally established that the time of integration for each integrator  203  should be from 20 to 200 msec for reliable detection of echo signals. The voltage readouts from all of the correlator channels form a signal representing the cross-correlation function. This signal has the form of a short radio impulse with the center frequency equal to the average frequency of the echo signal spectrum. In the above example, at a frequency band of 8-16 MHz, the center frequency is equal (8+16)/2=12 MHz. In computer  196  ( FIG. 12 ), the selection of the signal envelope is realized and comparison of the received maximum value with a chosen detection threshold is carried out. 
     In  FIG. 14 , a typical form for the signal envelope at the output of correlator  194  is shown. The signal shown in  FIG. 14  has a maximal amplitude in channel K, i.e. for the example above with the variable time delay set at 95 micro seconds—the measured value of time delay of the echo signal is equal 95+KTo=95+K 0.025 micro seconds. 
     After the echo signal is detected, the maximum amplitude values of the cross-correlation function envelope are accumulated, which correspond to measured time delay (corresponds to correlator channel K). These values are formed in computer  196  after reception of data from correlator  194 , with the time intervals equal to the integration time in integrator  203 . If an encoder  12  (or probe  42 ) is installed between a CPD source and the point of echo signal reception, then the signal of the maximum amplitude at the correlator output will have amplitude modulation which frequency corresponds to the ID of the encoder or probe (in the case of frequency division IDs). The remaining processing of the signal of the maximum amplitude cross-correlation function in computer  196  consists of defining the AM frequency for identification of encoder  12  (or probe  42 ) (i.e., decoding), which can be carried out by the ordinary spectral analysis method. The required accumulation time of signal readouts is selected the same way as is performed in the case of the ingress analysis defined previously. 
     U.S. Published Application No. US 2006/0248564 A1, published on Nov. 2, 2006, and U.S. Published Application No. US 2004/0245995 A1, published on Dec. 9, 2004, are incorporated herein by reference. 
     While the preferred embodiments of the invention have been particularly described in the specification and illustrated in the drawing, it should be understood that the invention is not so limited. Many modifications, equivalents and adaptations of the invention will become apparent to those skilled in the art without departing from the spirit and scope of the invention, as defined in the appended claims.