Abstract:
A PLL circuit includes a phase comparator for outputting a frequency control signal based on a result of comparison between phases of an input reference clock signal and a fed-back oscillation signal; an oscillation circuit, connected to the phase comparator, for outputting an oscillation signal having a frequency in accordance with the frequency control signal, a power source voltage, and a predetermined reference voltage; and a bias control circuit, connected to the oscillation circuit, for increasing either the potential difference between the reference voltage of the oscillation circuit and a ground voltage or the potential difference between the power source voltage of the oscillation circuit and the ground voltage. A transistor in the oscillation circuit can operate in a saturation area, thereby operating the PLL circuit at a high speed with a low power source voltage, without being easily affected by a variation in the temperature or other process conditions.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to a technique for operating a PLL (phase-locked loop) circuit at a high speed with a low power source voltage. 
     Priority is claimed on Japanese Patent Application No. 2007-042229, filed Feb. 22, 2007, the contents of which are incorporated herein by reference. 
     2. Description of the Related Art 
     The PLL circuit has a function of multiplying the frequency of a clock signal, controlling the skew with respect to a clock signal, or the like, and thus is an important circuit used in a semiconductor integrated circuit or an electronic device. With the advance of a low power source voltage and a high-speed operation of recent electronic devices, the PLL circuit should also operate with a low power source voltage and at a high speed. 
     For example, such a low power source voltage and a high-speed operation have been possible with respect to a DRAM (dynamic random access memory), which causes a reduction of the margin provided for the operation of a PLL circuit, which is included in the DRAM so as to control the relevant parts of the DRAM. 
     As a PLL circuit coping with such a low power source voltage, Japanese Unexamined Patent Application, First Publication No. H08-130465 discloses a PLL circuit having a wide control-voltage range with respect to a charge pump circuit. 
     Generally, the charge pump circuit is a main circuit which restricts the low power source voltage and the high-speed operation. 
       FIG. 13  is a circuit diagram showing a charge pump circuit and a loop filter which are generally used in a PLL circuit. In a charge pump circuit  1100 , a control voltage Vcont is (i) increased by outputting a current Iup to a loop filter  103  in accordance with an increase signal “UP” (i.e., signal for designating an increase in the control voltage), and (ii) decreased by receiving a current Idown from the loop filter  103  in accordance with a decrease signal “DOWN”. 
     Additionally, in the PLL circuit, a voltage-controlled oscillator (VCO) is connected to the control voltage Vcont so that the oscillation frequency of the VCO is controlled by the control voltage Vcont. Generally, in such a VCO, the higher the control voltage Vcont, the higher the oscillation frequency. 
     That is, the control voltage Vcont should be high so as to obtain a high oscillation frequency of the VCO. However, as a result, the voltage between the drain and the source (i.e., drain-source voltage Vds) of a PMOS transistor  1101 , which forms the charge pump circuit  1100 , should be low. 
     The operation of the charge pump circuit  11  in this process will be explained with reference to  FIG. 14 , which shows the dependency of the outflow current Iup (which flows by means of the PMOS transistor  1101 ) and the inflow current Idown (which flows by means of an NMOS transistor  1102 ) on the control voltage Vcont. As shown in  FIG. 14 , when the control voltage Vcont increases, the operating point of the PMOS transistor  1101  may shift from the point B, which belongs to a saturation area, to the point A, which belongs to a linear area (i.e., area  1200  which is close to the power source voltage VDD). 
     In the vicinity of the point B, when the control voltage Vcont varies, a small variation occurs in the outflow current Iup and the inflow current Idown. In contrast, in the vicinity of the point A, when the control voltage Vcont varies, there is a considerable variation in the outflow current Iup. 
     That is, in order that the VCO oscillates at a constant frequency, the control voltage Vcont varies depending on a variation in the temperature or other conditions with regard to the relevant processes, and the outflow current Iup considerably varies in accordance with the variation in the control voltage Vcont, thereby increasing the difference between the outflow current Iup and the inflow current Idown. 
     On the other hand, when decreasing the power source voltage VDD, the drain-source voltage Vds of the PMOS transistor  1101  (which forms the charge pump circuit) also decreases, similar to the above process. As a result, the PMOS transistor  1101  operates at the point A which belongs to the linear area, which also causes the above-described problem. 
     That is, when decreasing the power source voltage VDD by increasing the oscillation frequency of the PLL circuit, there occurs a difference between the values of the outflow current Iup and the inflow current Idown. In addition, there occurs a large variation in the outflow current Iup in accordance with a variation in the temperature or other process conditions. Therefore, even when the PLL circuit is locked, there occurs a phase difference between the signals output from the relevant phase comparator (i.e., the signals UP and DOWN), which causes a drift in the oscillation frequency of the VCO (i.e., depending on a variation in the temperature or other process conditions). 
     SUMMARY OF THE INVENTION 
     In light of the above circumstances, an object of the present invention is to provide a PLL circuit, which operates with a low power source voltage and at a high speed without being easily affected by a variation in the temperature or other process conditions. 
     Therefore, the present invention provides a PLL circuit comprising: 
     a phase comparator for outputting a frequency control signal based on a result of comparison between phases of an input reference clock signal and a fed-back oscillation signal; 
     an oscillation circuit, connected to the phase comparator, for outputting an oscillation signal having a frequency in accordance with the frequency control signal output from the phase comparator, a power source voltage, and a predetermined reference voltage; and 
     a bias control circuit, connected to the oscillation circuit, for increasing either the potential difference between the reference voltage of the oscillation circuit and a ground voltage or the potential difference between the power source voltage of the oscillation circuit and the ground voltage. 
     In accordance with the above structure, either the potential difference between the reference voltage of the oscillation circuit and a ground voltage or the potential difference between the power source voltage of the oscillation circuit and the ground voltage can be increased so that a transistor provided in the oscillation circuit can operate in a saturation area, thereby operating the PLL circuit at a high speed with a low power source voltage, without being easily affected by a variation in the temperature or other process conditions. 
     In a first typical example: 
     the oscillation circuit includes:
         a charge pump circuit into which the frequency control signal is input;   a loop filter into which a current in accordance with the frequency control signal and the power source voltage is input from the charge pump circuit, so as to output a control voltage; and   a voltage-controlled oscillator for outputting the oscillation signal having an oscillation frequency in accordance with the potential difference between the reference voltage and the control voltage output from the loop filter; and       

     the bias control circuit inputs the reference voltage into the voltage-controlled oscillator via a reference-voltage terminal thereof, where the input reference voltage is lower than the ground voltage. 
     In this case, as the reference voltage of the voltage-controlled oscillator is a negative voltage, the control voltage can be decreased, so that a (PMOS) transistor in the charge pump circuit for outputting the relevant current can operate in a saturation area. Accordingly, the outflow current and the inflow current with respect to the charge pump circuit can be almost identical to each other, thereby operating the PLL circuit at a high speed with a low power source voltage, without being easily affected by a variation in the temperature or other process conditions. 
     In a second typical example: 
     the oscillation circuit includes:
         a charge pump circuit into which the frequency control signal is input;   a loop filter into which a current in accordance with the frequency control signal and the power source voltage is input from the charge pump circuit, so as to output a control voltage; and   a voltage-controlled oscillator for outputting the oscillation signal having an oscillation frequency in accordance with the potential difference between the reference voltage and the control voltage output from the loop filter; and       

     the bias control circuit inputs the power source voltage into the charge pump circuit via a power source voltage terminal thereof, where the input power source voltage is higher than an original power source voltage. 
     In this case, as the power source voltage input into the power source voltage terminal of the charge pump circuit is higher than the original power source voltage (i.e., supplied to structural elements other than the charge pump circuit in the PLL circuit) supplied from the power source, a (PMOS) transistor in the charge pump circuit for outputting the relevant current can operate in a saturation area. Accordingly, the outflow current and the inflow current with respect to the charge pump circuit can be almost identical to each other, thereby operating the PLL circuit at a high speed with a low power source voltage, without being easily affected by a variation in the temperature or other process conditions. 
     In a preferable example of the first typical example, the bias control circuit is a negative-voltage generating circuit for inputting a negative voltage as the reference voltage input into the reference-voltage terminal of the voltage-controlled oscillator. 
     In a preferable example of the second typical example, the bias control circuit is a booster circuit for inputting a voltage obtained by boosting the original power source voltage, into the power source voltage terminal of the charge pump circuit. 
     In another preferable example of the first typical example,: 
     the bias control circuit includes:
         a negative-voltage generating circuit for generating a negative voltage; and   a first voltage control circuit for receiving the negative voltage via an input terminal thereof, changing the value of the negative voltage in accordance with a control signal input via a control terminal thereof, and inputting the changed negative voltage as the reference voltage into the reference-voltage terminal of the voltage-controlled oscillator.       

     In this case, as the value of the negative voltage can be controlled, the reference voltage of the voltage-controlled oscillator can have an appropriate value in accordance with the oscillation frequency. 
     In another preferable example of the second typical example,: 
     the bias control circuit includes:
         a booster circuit for boosting the original power source voltage; and   a second voltage control circuit for receiving the boosted power source voltage via an input terminal thereof, changing the value of the power source voltage in accordance with a control signal input via a control terminal thereof, and inputting the changed power source voltage into the power source voltage terminal of the charge pump circuit.       

     In this case, as the voltage higher than the original power source voltage can be controlled, the voltage for driving the charge pump circuit can have an appropriate value. 
     It is possible that the first voltage control circuit includes: 
     a first NMOS transistor whose gate and drain are connected to the control terminal; 
     a second NMOS transistor whose gate and drain are connected to the source of the first NMOS transistor; 
     a third NMOS transistor whose gate and drain are connected to the source of the second NMOS transistor, and whose source is connected to an output terminal used for outputting a signal to the voltage-controlled oscillator; 
     a fourth NMOS transistor whose drain is connected to the output terminal, and whose source is connected to the negative-voltage generating circuit; and 
     an amplifier, one input terminal of which is connected to the source of the second NMOS transistor and the gate and drain of the third NMOS transistor, and the other input terminal of which is grounded, wherein the output terminal of the amplifier is connected to the gate of the fourth NMOS transistor. 
     It is possible that the second voltage control circuit includes: 
     a first PMOS transistor whose source is connected to the booster circuit, and whose drain is connected to an output terminal used for outputting a signal to the charge pump circuit; and 
     an amplifier, one input terminal of which is connected to the control terminal, and the other input terminal is connected to the output terminal, wherein the output terminal of the amplifier is connected to the gate of the first PMOS transistor. 
     In another preferable example of the first typical example: 
     the voltage-controlled oscillator includes:
         a first PMOS transistor, a second PMOS transistor, and a third PMOS transistor, whose sources are connected to an input terminal used for receiving the control voltage from the loop filter; and   a first NMOS transistor, a second NMOS transistor, and a third NMOS transistor, whose sources are connected to the reference-voltage terminal;       

     the drains of the first PMOS transistor and the first NMOS transistor are connected to the gates of the second PMOS transistor and the second NMOS transistor; 
     the drains of the second PMOS transistor and the second NMOS transistor are connected to the gates of the third PMOS transistor and the third NMOS transistor; 
     the drains of the third PMOS transistor and the third NMOS transistor are connected to an output terminal used for outputting the oscillation signal; and 
     the gates of the first PMOS transistor and the first NMOS transistor are connected to the output terminal. 
     In another preferable example of the second typical example: 
     the frequency control signal is one of an increase signal for increasing the frequency of the oscillation signal and a decrease signal for decreasing the frequency of the oscillation signal; 
     the charge pump circuit includes:
         a first PMOS transistor and a second PMOS transistor, whose sources are connected to the power source voltage terminal, and whose gates are connected to a bias voltage terminal into which a bias voltage is input;   a third PMOS transistor, whose source is connected to the drain of the second PMOS transistor, wherein the increase signal is input into the gate of the third PMOS transistor,   a first NMOS transistor, whose drain and gate are connected to the drain of the first PMOS transistor, wherein the source of the first NMOS transistor is grounded;   a second NMOS transistor in which the decrease is input into the gate thereof, and the drain of the second NMOS transistor is connected to the drain of the third PMOS transistor; and   a third NMOS transistor, whose source is grounded, and whose drain and gate are respectively connected to the source of the second NMOS transistor and the gate of the first NMOS transistor; and       

     the drains of the third PMOS transistor and the second NMOS transistor are connected to an output terminal used for outputting a signal to the loop filter. 
     The present invention also provides a DRAM comprising: 
     a phase comparator for outputting a frequency control signal based on a result of comparison between phases of an input reference clock signal and a fed-back oscillation signal; 
     a charge pump circuit into which the frequency control signal is input; 
     a loop filter into which a current in accordance with the frequency control signal and the power source voltage is input from the charge pump circuit, so as to output a control voltage; 
     a voltage-controlled oscillator for outputting the oscillation signal having an oscillation frequency in accordance with the potential difference between the reference voltage and the control voltage output from the loop filter; 
     a negative-voltage generating part for generating a back bias voltage which is a negative voltage; and 
     a first voltage control circuit for receiving the back bias voltage via an input terminal thereof, changing the value of the back bias voltage in accordance with a control signal input via a control terminal thereof, and inputting the changed back bias voltage as the reference voltage into the voltage-controlled oscillator. 
     In accordance with the above structure, the negative voltage (i.e., the back bias voltage) of the DRAM is used as the reference voltage of the voltage-controlled oscillator. Therefore, the control voltage can be decreased, so that a (PMOS) transistor for outputting the current in the charge pump circuit can operate in a saturation area, without adding another specific circuit. Accordingly, the outflow current and the inflow current with respect to the charge pump circuit can be almost identical to each other, thereby operating the PLL circuit at a high speed with a low power source voltage, without being easily affected by a variation in the temperature or other process conditions. 
     The present invention also provides a DRAM comprising: 
     a phase comparator for outputting a frequency control signal based on a result of comparison between phases of an input reference clock signal and a fed-back oscillation signal; 
     a charge pump circuit into which the frequency control signal is input; 
     a loop filter into which a current in accordance with the frequency control signal and the power source voltage is input from the charge pump circuit, so as to output a control voltage; 
     a voltage-controlled oscillator for outputting the oscillation signal having an oscillation frequency in accordance with the potential difference between the reference voltage and the control voltage output from the loop filter; 
     a high-voltage generating part for outputting a word-line voltage higher than an original power source voltage; and 
     a second voltage control circuit for receiving the word-line voltage via an input terminal thereof, changing the value of the word-line voltage in accordance with a control signal input via a control terminal thereof, and inputting the changed word-line voltage into the charge pump circuit. 
     In accordance with the above structure, the charge pump circuit is driven by a voltage (i.e., the word-line voltage) higher than the voltage used for driving other circuits included in the DRAM, so that a (PMOS) transistor for outputting the current in the charge pump circuit can operate in a saturation area, without adding another specific circuit. Accordingly, the outflow current and the inflow current with respect to the charge pump circuit can be almost identical to each other, thereby operating the PLL circuit at a high speed with a low power source voltage, without being easily affected by a variation in the temperature or other process conditions. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a diagram showing the structure of a PLL circuit as a first embodiment of the present invention. 
         FIG. 2  is a circuit diagram showing the VCO of the first embodiment. 
         FIG. 3  is a graph showing the relationship between the charge pump current and the control voltage Vcont of the PLL circuit in the first embodiment. 
         FIG. 4  is a diagram showing the structure of a PLL circuit as a second embodiment of the present invention. 
         FIG. 5  is a circuit diagram showing the charge pump circuit and the loop filter of the second embodiment. 
         FIGS. 6A and 6B  are graphs showing the relationship between the charge pump current and the control voltage Vcont of the PLL circuit in the second embodiment. 
         FIG. 7  is a diagram showing the structure of a PLL circuit as a third embodiment of the present invention. 
         FIG. 8  is a circuit diagram showing the voltage control circuit of the third embodiment. 
         FIG. 9  is a diagram showing the structure of a PLL circuit as a fourth embodiment of the present invention. 
         FIG. 10  is a circuit diagram showing the voltage control circuit of the fourth embodiment. 
         FIG. 11  is a block diagram showing a DRAM and a PLL circuit included in the DRAM of a fifth embodiment of the present invention. 
         FIG. 12  is a block diagram showing a DRAM and a PLL circuit included in the DRAM of a sixth embodiment of the present invention. 
         FIG. 13  is a circuit diagram showing a charge pump circuit and a loop filter which are generally used in a conventional PLL circuit. 
         FIG. 14  is a diagram showing the dependency of the outflow current Iup and the inflow current Idown on the control voltage Vcont, with respect to the charge pump circuit in  FIG. 13 . 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Hereinafter, embodiments of the present invention will be described with reference to the appended figures. 
     First Embodiment 
     A first embodiment of the present invention will be explained with reference to  FIGS. 1 to 3 . 
       FIG. 1  is a diagram showing the structure of a PLL circuit as the first embodiment of the present invention. The present PLL circuit is of a charge pump type. 
     In  FIG. 1 , reference numeral  100  indicates a voltage-controlled oscillator (i.e., VCO), reference numeral  101  indicates a phase comparator, reference numeral  102  indicates a charge pump circuit, reference numeral  103  indicates a loop filter, reference numeral  104  indicates a capacitor (or capacitance) which forms the loop filter  103 , and reference numeral  105  indicates a negative-voltage generating circuit. 
     To the phase comparator  101 , a reference clock signal Vin and a fed-back oscillation signal are input, where the latter is an oscillation signal Vout of the VCO  100 , which is fed back. The increase signal UP and the decrease signal DOWN output from the phase comparator  101  are each input into the charge pump circuit  102 , as a frequency control signal used for controlling the oscillation frequency of a signal output from the VCO  100 . In addition, the output (i.e., for outputting the control voltage Vcont) of the charge pump circuit  102  is connected to one end of the capacitor  104  which forms the loop filter  103 , and the control voltage Vcont is also input into the VCO  100 . The other end of the capacitor  104  is grounded. Additionally, the signal (i.e., negative voltage Vb) output from the negative-voltage generating circuit  105  is input as the reference voltage into the VCO  100 . 
     The operation of the present PLL circuit will be explained below. First, the phase comparator  101  compares the phases of the reference clock signal and the fed-back oscillation signal, which are input into the pair of the inputs of the phase comparator  101 . Based on the result of the comparison, the phase comparator  101  outputs the increase signal UP or the decrease signal DOWN. For example, (i) when the frequency of the fed-back oscillation signal is lower than that of the reference clock signal, the phase comparator  101  outputs the increase signal UP, and (ii) when the frequency of the fed-back oscillation signal is higher than that of the reference clock signal, the phase comparator  101  outputs the decrease signal DOWN. 
     Next, based on the increase signal UP or the decrease signal DOWN, the charge pump circuit  102  outputs or receives the relevant current. 
     The loop filter  103  converts the current output from the charge pump circuit  102  into the control voltage Vcont, and also performs a filtering process. The VCO  100  generates an oscillation signal Vout in accordance with the control voltage Vcont, and outputs the signal, not only to the outside thereof, but also to the phase comparator  101  as the fed-back oscillation signal. The PLL circuit repeatedly performs the above-described operation, so as to output the oscillation signal Vout having a desired frequency which is locked. 
     The negative-voltage generating circuit  105  generates the negative voltage Vb which is lower than the ground voltage. The negative voltage Vb is input as a reference voltage from the reference-voltage terminal of the VCO  100 . Accordingly, the oscillation frequency of the VCO  100  is determined in accordance with the difference between the negative voltage Vb and the control voltage Vcont. 
     A concrete example of the VCO  100  will be shown with reference to  FIG. 2 . That is,  FIG. 2  is a circuit diagram showing the VCO  100  of the first embodiment. 
     In  FIG. 2 , reference numerals  201 ,  203 , and  205  indicate PMOS transistors, and reference numerals  202 ,  204 , and  206  indicate NMOS transistors. 
     With respect to the PMOS transistor  201  and the NMOS transistor  202 , the gates are connected to each other, and the drains are also connected to each other. 
     With respect to the PMOS transistor  203  and the NMOS transistor  204 , the drains are connected to each other, and the gates are also connected to each other, which are also connected to the drains of the PMOS transistor  201  and the NMOS transistor  202 . 
     With respect to the PMOS transistor  205  and the NMOS transistor  206 , the gates are connected to each other, and are also connected to the drains of the PMOS transistor  203  and the NMOS transistor  204 . In addition, the drains of the PMOS transistor  205  and the NMOS transistor  206  are connected to each other, and they output the oscillation signal Vout. They are also connected to the gates of the PMOS transistor  201  and the NMOS transistor  202 . 
     The sources of the PMOS transistors  201 ,  203 , and  205  are connected to each other, and the control voltage Vcont is input into the sources. In addition, the sources of the NMOS transistors  202 ,  204 , and  206  are connected to each other, and are also connected to the reference-voltage terminal, to which the negative voltage Vb is input as the reference voltage. 
     As described above, the negative voltage Vb is generated using the negative-voltage generating circuit  105 , and is input as the reference voltage of the VCO  100 , thereby decreasing the control voltage Vcont by the negative voltage Vb, without changing the oscillation frequency. 
     For example, if the oscillation frequency of the VCO  100  is 1.6 GHz when the reference voltage is 0 V and the control voltage Vcont is 1.2 V, then by using the negative voltage Vb which is −0.5 V as the reference voltage, the oscillation frequency of the VCO  100  becomes 1.6 GHz when the control voltage Vcont is approximately 0.7 V. 
       FIG. 3  is a graph showing the relationship between the charge pump current and the control voltage Vcont of the PLL circuit. As shown in  FIG. 3 , when the reference voltage of the VCO  100  is 0 V, the control voltage Vcont is present at the point A. However, when using the negative voltage Vb as the reference voltage, the control voltage Vcont shifts to the point B, thereby avoiding a linear area  300 , and also setting the outflow current Iup and the inflow current Idown to almost identical values. 
     That is, using the negative-voltage generating circuit  105  makes the transistor, by which the charge pump circuit  102  outputs the outflow current, operate in a saturation area. 
     Second Embodiment 
     A second embodiment of the present invention will be explained with reference to  FIGS. 4 to 6 .  FIG. 4  is a diagram showing the structure of a PLL circuit of the present embodiment. The present PLL circuit is also of the charge pump type. 
     In  FIG. 4 , reference numeral  400  indicates a VCO, reference numeral  402  indicates a charge pump circuit, and reference numeral  405  indicates a booster circuit. The other structural elements are identical to the corresponding elements in  FIG. 1 , and explanations thereof are omitted here. 
     To the phase comparator  101 , the reference clock signal Vin and a fed-back oscillation signal are input, where the latter is an oscillation signal Vout of the VCO  400 , which is fed back. The increase signal UP and the decrease signal DOWN output from the phase comparator  101  are each input into the charge pump circuit  402 . In addition, the output (i.e., for outputting the control voltage Vcont) of the charge pump circuit  402  is connected to one end of the capacitor  104  which forms the loop filter  103 , and the control voltage Vcont is also input into the VCO  400 . The other end of the capacitor  104  is grounded. Additionally, the signal (i.e., voltage Vp) output from the booster circuit  405  is input as the power source voltage into the source terminal of the charge pump circuit  402 . 
     The basic operation of the present PLL circuit is identical to that of the first embodiment, and explanations thereof are omitted. Below, distinctive operations of the present embodiment will be described. 
     The booster circuit  405  generates the voltage Vp, which is higher than the power source voltage VDD supplied from a power source. The charge pump circuit  402  is driven by the voltage Vp generated by the booster circuit  405 . That is, the charge pump circuit  402  operates by the power source voltage Vp (&gt;VDD), and the structural elements other than the charge pump circuit  402  operate by the power source voltage VDD. 
     A concrete example of the charge pump circuit  402  will be shown with reference to  FIG. 5 . That is,  FIG. 5  is a circuit diagram showing the charge pump circuit and the loop filter of the present embodiment. 
     In  FIG. 5 , reference numerals  501 ,  503 , and  504  indicate PMOS transistors, and reference numerals  502 ,  505 , and  506  indicate NMOS transistors. The other structural elements are identical to the corresponding elements in  FIG. 4 , and explanations thereof are omitted here. 
     With respect to the PMOS transistors  501  and  503 , the sources are connected to each other, and the gates are also connected to each other, to which a bias voltage Vbias is input. 
     The drain of the PMOS transistor  501  is connected to the source of the NMOS transistor  502 . The gate of the NMOS transistor  502  is connected to the drain thereof, and also connected to the gate of the NMOS transistor  506 . The sources of the NMOS transistors  502  and  506  are each grounded. 
     The source of the PMOS transistor  504  is connected to the drain of the PMOS transistor  503 , and the drain of the PMOS transistor  504  is connected to the drain of the NMOS transistor  505 , which is also connected to the loop filter  103  (i.e., the control voltage Vcont). The increase signal UP is input into the gate of the PMOS transistor  504 . 
     The source of the NMOS transistor  505  is connected to the drain of the NMOS transistor  506 , and the decrease signal DOWN is input into the gate of the NMOS transistor  505 . 
     The values of the outflow current Iup and the inflow current Idown are determined in accordance with the specific bias voltage Vbias. 
     In the present charge pump circuit  402 , when the increase signal UP is input, the PMOS transistor  504  is switched on, so that the outflow current Iup is output to the loop filter  103  via the PMOS transistor  503 . On the other hand, when the decrease signal DOWN is input, the NMOS transistor  505  is switched on, so that the inflow current Idown is drawn from the loop filter  103  via the NMOS transistor  506 . 
     Accordingly, the charge pump circuit  402  controls the control voltage Vcont. 
     As described above, the voltage Vp, which is higher than the power source voltage VDD, is generated using the booster circuit  405 , and it functions as the power source voltage used for driving the charge pump circuit  402 . Therefore, it is possible to increase the drain-source voltage Vds of the PMOS transistor  503 . 
     For example, if the drain-source voltage Vds of the PMOS transistor  503  is approximately 0.6 V when the power source voltage Vp and the control voltage Vcont of the charge pump circuit  402  are respectively 1.8 V and 1.2 V, then by setting the power source voltage Vp to 3 V, the drain-source voltage Vds of the PMOS transistor  503  becomes approximately 1.8 V. 
       FIGS. 6A and 6B  are graphs showing the relationships between the charge pump current and the control voltage Vcont of the present PLL circuit. As shown in  FIG. 6A , when the power source voltage is VDD, the control voltage Vcont is present at the point A which belongs to a linear area  600 , where there is a considerable difference between the outflow current Iup and the inflow current Idown. However, as shown in  FIG. 6B , when the power source voltage is Vp which is larger than VDD, the control voltage Vcont is also present at the point A, but it is possible to avoid a linear area  601 , and an area can be used where the outflow current Iup and the inflow current Idown are almost identical to each other. 
     That is, using the booster circuit  405  allows the PMOS transistor  503 , by which the charge pump circuit  402  outputs the outflow current, to operate in a saturation area. 
     When a circuit consisting of the charge pump circuit  102 , the loop filter  103 , and the VCO  100  of the first embodiment and a circuit consisting of the charge pump circuit  402  , the loop filter  103 , and the VCO  400  are each regarded as an oscillation circuit, the negative-voltage generating circuit  105  of the first embodiment and the booster circuit  405  of the second embodiment each function as a bias control circuit for supplying a negative voltage or a voltage higher than the power source voltage to the oscillation circuit. 
     Third Embodiment 
     A third embodiment of the present invention will be explained with reference to  FIGS. 7 and 8 . In  FIG. 7 , reference numeral  700  indicates a voltage control circuit (i.e., negative-voltage control part). The other structural elements are identical to the corresponding elements in  FIG. 1 , and explanations thereof are omitted here. 
     To the phase comparator  101 , the reference clock signal Vin and the fed-back oscillation signal are input, where the latter is the oscillation signal Vout of the VCO  100 , which is fed back. The increase signal UP and the decrease signal DOWN output from the phase comparator  10  are each input into the charge pump circuit  102 . In addition, the output (i.e., for outputting the control voltage Vcont) of the charge pump circuit  102  is connected to one end of the capacitor  104  which forms the loop filter  103 , and the control voltage Vcont is also input into the VCO  100 . The other end of the capacitor  104  is grounded. Additionally, the signal (i.e., negative voltage VB) output from the negative-voltage generating circuit  105  is input into the voltage control circuit  700 . In addition, the signal (i.e., negative voltage Vb) output from the voltage control circuit  700  is input as the reference voltage into the VCO  100 . Furthermore, a programmable voltage Vprog is input into the voltage control circuit  700  via a control terminal thereof. 
     The present PLL circuit is also of the charge pump type. 
     The basic operation of the present PLL circuit is identical to that of the first embodiment, and explanations thereof are omitted. Below, distinctive operations of the present embodiment will be described. 
     The negative-voltage generating circuit  105  generates the negative voltage VB. The voltage control circuit  700  controls the negative voltage VB in accordance with the programmable voltage Vprog, so as to convert the negative voltage VB into the negative voltage Vb. The negative voltage Vb is input as the reference voltage into the VCO  100 . Accordingly, the oscillation frequency of the VCO  100  is determined in accordance with the difference between the negative voltage Vb and the control voltage Vcont. 
     A concrete example of the voltage control circuit  700  will be shown with reference to  FIG. 8 . That is,  FIG. 8  is a circuit diagram of the voltage control circuit  700 . 
     In  FIG. 8 , reference numeral  800  indicates an amplifier, and reference numerals  801  to  804  indicate NMOS transistors. 
     The gate and the drain of the NMOS transistor  801  are connected to each other, and are also connected to the control terminal, into which the programmable voltage Vprog is input. 
     The gate and the drain of the NMOS transistor  802  are connected to each other, and are also connected to the source of the NMOS transistor  801 . 
     The gate and the drain of the NMOS transistor  803  are connected to each other, and are also connected to the source of the NMOS transistor  802  and one of two input terminals of the amplifier  800 . 
     The other input terminal of the amplifier  800  is grounded, and the output terminal thereof is connected to the gate of the NMOS transistor  804 . The drain of the NMOS transistor  804  is connected to the source of the NMOS transistor  803 , and also outputs the negative voltage Vb. In addition, the negative voltage VB is input into the source of the NMOS transistor  804 . 
     In this example, the NMOS transistors  801  to  803  have the same transistor size. 
     The operation of the voltage control circuit  700  will be explained below. Here, it is assumed that the programmable voltage Vprog input from the control terminal is 0.8 V, and the negative voltage VB is −0.5 V. As the above-described other input terminal of the amplifier  800  is grounded, the above-described one input terminal thereof is 0 V. That is, the gate voltage and the drain voltage of the NMOS transistor  803  are each 0 V. Therefore, a current in accordance with the programmable voltage Vprog and the resistances of the NMOS transistors  801  and  802  flows through the NMOS transistors  801  to  803 , and the current further flows through the NMOS transistor  804 . 
     As the NMOS transistors  801  to  803  have the same transistor size, the voltage drop generated through the NMOS transistors  801  and  802  is 0.8 V, and the voltage drop generated through the NMOS transistor  803  is 0.4 V. That is, the negative voltage Vb is −0.4 V. 
     Similarly, the negative voltage Vb can be changed by changing the programmable voltage Vprog. For example, when the programmable voltage Vprog is 0.7 V, the negative voltage Vb is −0.35 V, and when the programmable voltage Vprog is 0.6 V, the negative voltage Vb becomes −0.3 V. 
     In accordance with the above-described circuit, the reference voltage of the VCO  100  can be programmably controlled and set to an appropriate value in accordance with the oscillation frequency. 
     That is, similar to the first embodiment, with reference to  FIG. 3  which shows the relationship between the charge pump current and the control voltage Vcont, when the reference voltage of the VCO  100  is 0 V, the control voltage Vcont is present at the point A. However, when using the negative voltage Vb as the reference voltage, the control voltage Vcont shifts to the point B, so that the linear area  300  can be avoided, and an area can be used where the outflow current Iup and the inflow current Idown are almost identical to each other. 
     That is, using the negative-voltage generating circuit  105  and the voltage control circuit  700  allows the transistor, by which the charge pump circuit  102  outputs the outflow current, to operate in a saturation area. Here, the first voltage control circuit of the present invention corresponds to the voltage control circuit  700 . In addition, the NMOS transistors  801  to  803  may have different transistor sizes. 
     Fourth Embodiment 
     A fourth embodiment of the present invention will be explained with reference to  FIGS. 9 and 10 .  FIG. 4  is a diagram showing the structure of a PLL circuit of the present embodiment. In  FIG. 4 , reference numeral  900  indicates a voltage control circuit (i.e., voltage control part). The other structural elements are identical to the corresponding elements in  FIG. 4 , and explanations thereof are omitted here. 
     To the phase comparator  101 , the reference clock signal Vin and the fed-back oscillation signal are input, where the latter is the oscillation signal Vout of the VCO  400 , which is fed back. The increase signal UP and the decrease signal DOWN output from the phase comparator  101  are each input into the charge pump circuit  402 . In addition, the output (i.e., for outputting the control voltage Vcont) of the charge pump circuit  402  is connected to the one end of the capacitor  104  which forms the loop filter  103 , and the control voltage Vcont is also input into the VCO  400 . The other end of the capacitor  104  is grounded. Additionally, the signal (i.e., voltage VP) output from the booster circuit  405  is input into the voltage control circuit  900 . The signal (i.e., voltage Vp) output from the voltage control circuit  900  is input as the power source voltage into the charge pump circuit  402 . Furthermore, a programmable voltage Vprog is input into the voltage control circuit  900  via a control terminal thereof. 
     The present PLL circuit is also of the charge pump type. 
     The basic operation of the present PLL circuit is identical to that of the first embodiment, and explanations thereof are omitted. Below, distinctive operations of the present embodiment will be described. 
     The booster circuit  405  generates the voltage VP, which is higher than the power source voltage VDD. The voltage control circuit  900  controls the voltage VP in accordance with the programmable voltage Vprog, so as to convert the voltage VP into the voltage Vp. The charge pump circuit  402  is driven by the voltage Vp output from the voltage control circuit  900 . That is, the charge pump circuit  402  operates by the power source voltage Vp (&gt;VDD), and the elements other than the charge pump circuit  402  operate by the power source voltage VDD. 
     A concrete example of the voltage control circuit  900  will be shown with reference to  FIG. 10 . That is,  FIG. 10  is a circuit diagram of the voltage control circuit  900 . 
     In  FIG. 10 , reference numeral  1000  indicates an amplifier, and reference numeral  1001  indicates a PMOS transistor. 
     One input terminal of the amplifier  1000  functions as the control terminal of the voltage control circuit  900 , and the programmable voltage Vprog is input into this control terminal. The other input terminal of the amplifier  1000  is connected to the drain of the PMOS transistor  1001 , and is also used for outputting the voltage Vp. The output terminal of the amplifier  1000  is connected to the gate of the PMOS transistor  1001 . In addition, the voltage VP is input into the source of the PMOS transistor  1001 . 
     Next, the operation of the voltage control circuit  900  will be explained. First, it is assumed that the programmable voltage Vprog is 2.5 V, and the voltage VP is 3.0 V. As the programmable voltage Vprog (2.5V) is input into the above-described one input terminal of the amplifier  1000 , the voltage Vp at the other input terminal of the amplifier 1000 is also 2.5 V. 
     Similarly, the voltage Vp can be changed by changing the programmable voltage Vprog. For example, when the programmable voltage Vprog is 2.8 V, the voltage Vp is also 2.8 V. That is, in the present circuit, the power source voltage of the charge pump circuit  402  can be programmably controlled. 
     Therefore, similar to the second embodiment, in the present embodiment, the linear area  601  (see  FIG. 6B ) can be avoided as indicated by the point A, and an area can be used where the outflow current Iup and the inflow current Idown are almost identical to each other. 
     That is, using the booster circuit  405  and the voltage control circuit  900  allows the PMOS transistor  503 , by which the charge pump circuit  402  outputs the outflow current, to operate in a saturation area. In addition, the second voltage control circuit of the present invention corresponds to the voltage control circuit  900 . 
     Fifth Embodiment 
     A fifth embodiment of the present invention will be explained with reference to  FIG. 11 . The present embodiment relates to a PLL circuit which is included in a DRAM. 
       FIG. 11  is a block diagram showing the DRAM and the PLL circuit included in the DRAM of the fifth embodiment. 
     In  FIG. 11 , reference numeral  2000  indicates the DRAM, reference numeral  2100  indicates a high-voltage generating part  2100 , reference numeral  2200  indicates a negative-voltage generating part, and reference numeral  2300  indicates the PLL circuit. The other structural elements are identical to the corresponding elements in  FIG. 7 , and explanations thereof are omitted here. In addition, in the DRAM  2000 , structural elements other than those shown in  FIG. 11  are not shown in the figure. 
     The DRAM  2000  includes the PLL circuit  2300  used for controlling each part in the DRAM. The PLL circuit  2300  has a structure in which the negative-voltage generating circuit  105  is excluded from the PLL circuit of the third embodiment. 
     The DRAM  2000  also includes the negative-voltage generating part  2200  for generating a back bias voltage Vbb which is a negative voltage, and the high-voltage generating part  2100  for generating a word-line voltage Vpp which is higher than the power source voltage VDD. As generally known, the back bias voltage Vbb and the word-line voltage Vpp are voltages used for operations such as writing, reading, and deletion of data with respect to the DRAM  2000 . For example, regarding a DRAM which operates with the power source voltage VDD of 1.8 V, the back bias voltage Vbb is −0.5 V, and the word-line voltage Vpp is 3 V. 
     As described in the third embodiment, the PLL circuit in accordance with the present invention can be implemented by inputting the back bias voltage Vbb into the voltage control circuit  700 , thereby omitting the negative-voltage generating circuit  105 . That is, the reference voltage of the VCO  100  is the negative voltage provided by the DRAM  2000  which is controlled by the PLL circuit  2300 . 
     Similar to the first embodiment, in accordance with the structure of the fifth embodiment, when the reference voltage of the VCO  100  is 0 V, the control voltage Vcont is present at the point A (see  FIG. 3 ). However, when using the negative voltage Vb as the reference voltage, the control voltage Vcont shifts to the point B, thereby avoiding the linear area  300 , and also setting the outflow current Iup and the inflow current Idown to almost identical values. 
     Additionally, similar to the first embodiment, the back bias voltage Vbb may be directly applied as the reference voltage of the VCO  100 , without using the voltage control circuit  700 . 
     Sixth Embodiment 
     A sixth embodiment of the present invention will be explained with reference to  FIG. 12 . The present embodiment also relates to a PLL circuit which is included in a DRAM. 
       FIG. 12  is a block diagram showing the DRAM and the PLL circuit included in the DRAM of the sixth embodiment. 
     In  FIG. 12 , reference numeral  3000  indicates the DRAM, reference numeral  3100  indicates the PLL circuit. The other structural elements are identical to the corresponding elements in  FIGS. 9 and 11 , and explanations thereof are omitted here. In addition, in the DRAM  3000 , structural elements other than those shown in  FIG. 12  are not shown in the figure. 
     The DRAM  3000  includes the PLL circuit  3100  used for controlling each part in the DRAM. The PLL circuit  3100  has a structure in which the booster circuit  405  is excluded from the PLL circuit of the fourth embodiment. In addition, similar to the fifth embodiment, the DRAM  3000  also includes the negative-voltage generating part  2200  for generating the back bias voltage Vbb which is a negative voltage, and a high-voltage generating part  2100  for generating the word-line voltage Vpp which is higher than the power source voltage VDD. 
     In the present embodiment, the PLL circuit in accordance with the present invention can be implemented by inputting the word-line voltage Vpp into the voltage control circuit  900 , thereby omitting the booster circuit  405 . That is, the charge pump circuit  402  is driven by a voltage higher than the power source voltage provided by the DRAM  3000  which is controlled by the PLL circuit  3100 . 
     Similar to the second embodiment, in accordance with the structure of the sixth embodiment, as shown by the point A in  FIG. 6B , the linear area  601  can be avoided, and the outflow current Iup and the inflow current Idown can be set to almost identical values. 
     Additionally, similar to the second embodiment, the word-line voltage Vpp may be directly applied as the power supply voltage of the charge pump circuit  402  without using the voltage control circuit  900 , so that the charge pump circuit  402  is driven by the word-line voltage Vpp. 
     While preferred embodiments of the invention have been described and illustrated above, it should be understood that these are exemplary embodiments of the invention and are not to be considered as limiting. Additions, omissions, substitutions, and other modifications can be made without departing from the scope of the present invention. Accordingly, the invention is not to be considered as being limited by the foregoing description, and is only limited by the scope of the appended claims. 
     For example, the VCO and the charge pump circuit may each have a structure other than those described above. 
     The structure of the PLL circuit is also not limited to those described above. For example, (i) the signal output from the VCO may be frequency-divided using a frequency divider, and then input into the phase comparator, or (ii) the reference clock signal may be frequency-divided using a frequency divider, and then input into the phase comparator. 
     In addition, the negative voltages Vb and VB, the voltages Vp and VP, the back bias voltage Vbb, and the word-line voltage Vpp are provided as examples, and another kind of voltage may be used. 
     Furthermore, the PLL circuit may simultaneously include some structural elements belonging to different embodiments described above. 
     The oscillation circuit of the present invention corresponds to (i) a circuit consisting of the charge pump circuit  102 , the loop filter  103 , and the VCO  100  in the first embodiment and the third embodiment, and (ii) a circuit consisting of the charge pump circuit  402 , the loop filter  103 , and the VCO  400  in the second embodiment and the fourth embodiment. 
     The bias control circuit of the present invention corresponds to (i) the negative-voltage generating circuit  105  in the first embodiment, (ii) the booster circuit  405  in the second embodiment, (iii) a circuit consisting of the voltage control circuit  700  (i.e., the first voltage control circuit) and the negative-voltage generating circuit  105  in the third embodiment, and (iv) a circuit consisting of the voltage control circuit  900  (i.e., the second voltage control circuit) and the booster circuit  405  in the fourth embodiment. 
     Additionally, in the voltage-controlled oscillator of the present invention, the first PMOS transistor corresponds to the PMOS transistor  201 , the second PMOS transistor corresponds to the PMOS transistor  203 , the third PMOS transistor corresponds to the PMOS transistor  205 , the first NMOS transistor corresponds to the NMOS transistor  202 , the second NMOS transistor corresponds to the NMOS transistor  204 , and the third NMOS transistor corresponds to the NMOS transistor  206 . 
     On the other hand, in the charge pump circuit of the present invention, the first PMOS transistor corresponds to the PMOS transistor  501 , the second PMOS transistor corresponds to the PMOS transistor  503 , the third PMOS transistor corresponds to the PMOS transistor  504 , the first NMOS transistor corresponds to the NMOS transistor  502 , the second NMOS transistor corresponds to the NMOS transistor  505 , and the third NMOS transistor corresponds to the NMOS transistor  506 . 
     Additionally, in the first voltage control circuit of the present invention, the first NMOS transistor corresponds to the NMOS transistor  801 , the second NMOS transistor corresponds to the NMOS transistor  802 , the third NMOS transistor corresponds to the NMOS transistor  803 , the fourth NMOS transistor corresponds to the NMOS transistor  804 , and the amplifier corresponds to the amplifier  800 . 
     On the other hand, in the second voltage control circuit of the present invention, the first PMOS transistor corresponds to the PMOS transistor  1001 , and the amplifier corresponds to the amplifier  1000 .