Abstract:
A method for RF communication between transceivers in a radio frequency identification system that improves range, decreases multipath errors and reduces the effect of outside RF source interference by employing spread spectrum techniques. By pulse amplitude modulating a spread spectrum carrier before transmission, the receiver can be designed for simple AM detection, suppressing the spread spectrum carrier and recovering the original data pulse code waveform. The data pulse code waveform has been further encrypted by a direct sequence pseudo-random pulse code. This additional conditioning prevents the original carrier frequency components from appearing in the broadcast power spectra and provides the basis for the clock and transmit carrier of the transceiver aboard an RFID tag. Other advantages include high resolution ranging, hiding transmissions from eavesdroppers, and selective addressing.

Description:
CROSS REFERENCE TO RELATED APPLICATION 
     This application is a divisional of application Ser. No. 08/800,918, filed Feb. 13, 1997, now U.S. Pat. No. 5,974,078, which is a divisional of application Ser. No. 08/348,274, filed Nov. 30, 1994, now U.S. Pat. No. 5,825,806, issued Oct. 20, 1998, which is a continuation of application Ser. No. 08/032,384, filed Mar. 17, 1993, now U.S. Pat. No. 5,539,775, issued Jul. 23, 1996. 
    
    
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     This invention relates to data communication with low power radio frequency (RF) transceivers, specifically to data communication between RF identification (RFID) tags and an operator (interrogator) and, more particularly, to using spread spectrum techniques to simplify the tag receiver design, reduce tag cost, increase the range and reduce interference on the RF communication channel. 
     2. State of the Art 
     An RFID tag is a small radio transceiver that can be attached to a movable article to help keep track of its whereabouts and status. A small, lightweight, inexpensive tag is desirable. To miniaturize the tag, the size of the circuitry is reduced by using an integrated circuit (IC) design. The simpler the circuit, the smaller the resulting IC becomes, thereby reducing the cost. 
     Operational environment factors can disrupt the reliability of these weak communication links. Reflective and refractive properties of the environment introduce possible multipath errors, and outside RF sources introduce interference in the received signal. 
     SUMMARY OF THE INVENTION 
     The principal object of this invention is to provide a method by which an RFID tag can communicate more information over much greater distances through harsher RF environments than previously available, while reducing the size and cost of the tag IC. 
     The secondary objects of this invention are to provide: simple AM reception by the receiver circuit without having to use complex synchronization schemes, external frequency references such as quartz crystals and complex circuits like Costas loops; 
     interrogator transmission power to be less than or equal to 1 Watt, allowing unlicensed operation within FCC guidelines; 
     extension of the range from interrogator to RFID tag; 
     limitation of eavesdropping of signals by outside parties; and 
     selective addressing of particular tags or communicating with more than one tag simultaneously in any given transmission. 
     These and other objects are achieved by modulating a spectrally spread carrier with a pulse code waveform representing information pertaining to the article to which the tag is attached. 
     One advantage of using a spread spectrum modulated signal is the enhanced interference rejection obtained during the demodulation process. The effect is a signal to noise gain over traditional narrow band broadcasting techniques. 
     Using spread spectrum techniques gives good range with relatively little complexity on the tag. The high complexity is contained in the interrogator design. 
     Other advantages are: selectively addressing a particular receiver, pulse code multiplexing allowing addressing of a plurality of receivers in any given transmission, broadcast transmissions with low density power spectra for signal hiding, message encryption to discourage eavesdroppers, and high resolution ranging between transmitter and receiver. 
     Although this invention was specifically designed for RFID tag communication, there are applications for it in hand-held walkie-talkies, pagers, mobile phones, cordless telephones, cordless microphones and musical instruments, cordless computer network communication links, and intercoms. 
    
    
     BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS 
     FIG. 1 is a simplified drawing of the type of radio frequency identification system to which this invention applies. 
     FIG. 2 is a block diagram of the transmitter part of the interrogator. 
     FIG. 3 is a series of time domain plots representing binary phase shift keying (BPSK) the carrier with a direct sequence pseudo-random pulse code. 
     FIG. 4 is a frequency domain plot of an ideally spectrally spread carrier. 
     FIGS. 5A-5B are a realistic time domain plot and frequency domain plot of the BPSK spectrally spread carrier. 
     FIG. 6 is a series of time domain plots representing pulse amplitude modulating the spectrally spread carrier with a pulse code waveform. 
     FIGS. 7A-7B are a realistic time domain plot and frequency domain plot of a pulse amplitude modulated non-spectrally spread carrier. 
     FIGS. 8A-8B are a realistic time domain plot and frequency domain plot of a pulse amplitude modulated spectrally spread carrier. 
     FIG. 9 is a block diagram of the transceiver in the REMOTE RFID tag. 
     FIG. 10 is a time domain plot representing one data bit period of the received signal  1 +PN 1 (t)d(t) after the carrier has been removed. 
     FIG. 11 is a time domain plot representing one data bit period of the received signal without carrier that has been gated by PN 1 (t) in sync. 
     FIG. 12 is a time domain plot representing one data bit period of the received signal without carrier that has been gated by PN 1 (t) that is 3 chips out of sync. 
     FIG. 13 is a time domain plot representing one data bit period of the received signal without carrier that has been gated by the logical compliment of PN 1 (t) in sync when d(t)=1. 
     FIG. 14 is a time domain plot representing one data bit period of the received signal without carrier that has been gated by PN 1 (t) in sync when d(t)=1. 
     FIG. 15 is a time domain plot representing one data bit period of the received signal without carrier that has been gated by the logical compliment of PN 1 (t) in sync when d(t)=−1. 
     FIG. 16 is a time domain plot representing one data bit period of the received signal without carrier that has been gated by PN 1 (t) in sync when d(t)=−1. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Referring now to FIG. 1, a typical arrangement of an RFID system includes a plurality of operator controlled interrogators A that communicate via RF links with a plurality of RFID tags B which are attached to articles C located somewhere in the vicinity of the interrogators. 
     An interrogator transmits a signal to a particular tag requesting or updating the status of the article associated with that tag. Status information can include the article&#39;s name, owner, address, destination, pertinent dates, weight, etc. 
     The tag receives the signal and awakens from its power conserving quiescent sleep state. The tag interprets the request and decides on a response which is then transmitted back to the requesting interrogator. A receiver in the interrogator analyzes the return signal, determining the status of the article. Its location can be determined by triangulation with multiple interrogators or multiple antennas. This information is then displayed to the operator. 
     Referring now to FIG. 2, which is a block diagram of the transmitter part of the interrogator transceiver, a crystal controlled oscillator  1  generates a constant amplitude sinusoidal carrier with a frequency of 2441.75 MHZ. 
     This carrier is modulated in the balanced modulator  2  by a pseudo-random direct sequence pulse code, PN 2 , generated by the pseudo-noise (PN) generator  3 . PN 2  has a chip rate of about 40 Mega-chips per second. 
     The resulting spectrally spread carrier is then modulated  4  by another pulse code waveform generated by combining a data waveform  5  and another pseudo-noise waveform, PN 1 (t)  6 . The resulting signal to be transmitted is sent to a power amplifier  7 , and then to the antenna  8 . 
     The data waveform, d(t) represents information to be transmitted to the REMOTE and has a data rate of about 2 mega-bits per second. PN 1 (t) has a chip rate equal to or less than PN 2 . In this embodiment, PN 1 (t) and d(t) are multiplied together to form the modulating unipolar waveform,  1 +PN 1 (t)d(t). 
     The waveforms of FIG. 3 refer in detail to operation of the first balanced modulation of the carrier. Modulation occurs by binary phase shift keying (BPSK) the original constant amplitude carrier  9 , with the PN 2  pseudo-random direct sequence pulse code  10 . During the time when PN 2  is low, the carrier  11  is keyed with a 180 degree phase shift from when PN 2  is high. For illustration purposes, the sinusoidal carrier is shown at a much lower frequency than has been suggested for this embodiment. 
     The resulting waveform has a power spectrum which is ideally spread according to the (sin(x)/x)**2 function shown in FIG.  4 . The center of the main lobe is at the original carrier frequency Fc. The mainlobe bandwidth (null to null) is now twice the clock frequency of the PN 2  waveform Fg. A more realistic representation of the time domain signal and its power spectra appears in FIG. 5 a  and FIG. 5 b  respectively. 
     FIG. 6 illustrates in detail the operation of the modulation of the spectrally spread carrier by the pulse code waveform  1 +PN 1 (t)d(t). The spectrally spread carrier  11  of FIG. 3 is viewed over a much greater time period  12 . One must keep in mind that there are over 150 cycles of the original carrier for every bit (amount of time between possible changes, high-to-low or low-to-high) of the  1 +PN 1 (t)d(t) modulating waveform  13 . 
     Pulse amplitude modulation involves multiplying the spread spectrum carrier by the modulating waveform  14 . It is important to note that pulse amplitude modulating a non-spectrally spread carrier results in a power spectrum with an undesirable spike  15  at the carrier frequency as shown in FIG. 7 a  and FIG. 7 b . By spectrally spreading the carrier, the resultant power spectra is more acceptable as shown in FIG. 8 a  and FIG. 8 b.    
     Referring now to FIG. 9, the receiver portion of the remote RFID tag receives signals via an antenna  16 . In order to sense the existence of a signal meaningful to the REMOTE, the signals are sent to a lowpass filter  17 . This voltage is compared to a value Vo in a comparator  18 . If the voltage is greater, the comparator outputs a true, informing the sequential mode logic  19  that a signal is present. The receiver is then initialized, awakening from its quiescent sleep state and the PN 1 (t) waveform generation circuitry is set to acquisition mode through the PN epoch controller  20 . 
     In an awakened REMOTE, the signal arriving at the antenna  16  is sent to several stages of RF (S-band) amplification and filtering  21 . As much RF amplification as attainable is provided. The bandpass filter characteristics are a center frequency of the original carrier (2441.75 MHZ) and a bandwidth of around 84 MHZ, which corresponds to the main lobe of the transmitted signal. Although this bandpass filtering is shown as a single block  22 , the circuitry accomplishing this filtering will likely be distributed throughout the RF amplifier circuits. 
     The resulting signal is sent to a full wave envelope demodulator  23 , which recovers the amplitude modulating waveform as  1 +PN 1 (t)d(t). Some added lowpass filtering  24  and baseband amplification  25  further increases the voltage level of the waveform to a more useful value (approx. 0.5 Volts peak-to-peak). 
     As seen in FIG. 10, the waveform  1 +PN 1 (t)d(t) is essentially two-level in nature, exhibiting positive and negative transitions between the limits of 0 and Vs which correspond to the PN 1 (t) waveform, and a small amount of additive noise or interference. Here, PN 1 (t) is represented by the chip sequence: 1 1 1 −1 −1 1 −1 −1, and spans exactly one data bit period, Td. It can be assumed during this time period that d(t)=1; if it were a −1, the waveform would be inverted. In order to facilitate synchronization, d(t) is equal to 1 and thus non-transitioning during a preamble portion of the transmitted signal. 
     Referring back to FIG. 9, the transition pulse generator  26  acts to differentiate the 1+PN 1 (t)d(t) waveform, producing “spikes” at the transition times, which are then rectified (absolute value) to furnish a unipolar train of transition derived pulses. Spectrally, this waveform is rich in the PN 1 (t) code clock frequency. It&#39;s injected into a locked multivibrator  27  (or a phase-locked loop oscillator) which produces a synchronized pulse for every chip of the received signal. In effect, the waveform produced is a relatively stable PN 1  clock waveform used by the PN 1  generator  28 . Once synchronization to the incoming PN 1  sequence is achieved, the locked multivibrator  27  is used as the tag&#39;s clock for processing the incoming data and commands. If a return transmission is requested, the locked multivibrator  27  becomes the source for the tag transmitter carrier frequency. 
     The PN 1  generator  28 , running from the PN-clock waveform, generates the tag&#39;s version of the PN 1 (t) chip sequence. This will initially be time offset by an integral number of chips (n) from the sequence that makes up the received waveform 1+PN 1 (t)d(t). The PN Sync Detector  29  eliminates this offset. 
     The PN Sync Detector accomplishes its objective by doing a sequential correlation comparison. A MOSFET switch  30  gates the 1+PN 1 (t)d(t) code received using the PN 1  code generated. If the two codes are in sync, the output of the gate is a waveform, as shown in FIG. 11, having an average value around Vs/2. If the two codes are unaligned, say offset by three chips, the gate output is a waveform, as shown in FIG. 12, having an average value around Vs/4 or less. By usual spread spectrum standards, correlation discrimination by a factor of two would be considered poor, but under very high signal to noise ratio (SNR) conditions, which are present in the forward link, it is acceptable. 
     The proper, in-sync starting point for the PN 1  code sequence is discovered by generating PN 1  at all possible starting points and recording the results. Each starting point represents a waveform, which is a possible candidate for being the correct in-sync PN 1  waveform. The candidate having the largest correlation value will be the one used. A more detailed explanation of this process follows. 
     Upon initialization, the Sequential Mode Logic  19 , in conjunction with the timing events T 1  and T 2   31 , initializes the sample-and-hold (S&amp;H) memories  32  and  33 . T 2  occurs later than T 1 . The event of T 1  causes the output of LPF 4   34  to be stored in S&amp;H # 1  as value E 1 . Likewise, T 2  results in E 1  being stored in S&amp;H # 2  as value E 2 . LPF 4   34  is responsible for smoothing the correlation voltage, producing a more accurate representation of the correlation than what is available from the broader LPF 2   35 , which is matched to the bit period. Initially, the sequential mode logic sets the PN epoch controller  20  to produce a shift-register word which is loaded into the PN generator&#39;s shift-register directing the generator to produce the PN 1  code sequence at a particular starting point called epoch- 1 . If the code sequence is made up of L chips, there are L possible starting points and L possible epochs. (In the example waveform in FIG. 4, L=8.) The PN sync detector must find epoch-n corresponding to the largest correlation, with 1&lt;=n&lt;=L. Once the generator is producing PN 1  based on epoch- 1 , the correlation value output from LPF 4  is transferred to S&amp;H # 1  at T 1  and then to S&amp;H # 2  at T 2 . 
     Next, the chip sequence based on epoch- 2  is generated. The output from LPF 4  is again sent to S&amp;H # 1  at T 1 . If E 1  is greater than E 2 , the correlation at epoch- 2  is greater than the correlation at epoch- 1 . This is communicated by the output of the comparator  36  to the sequential mode logic which transfers E 1  to S&amp;H # 2  and makes note of the epoch value n (in this case 2) for which the transfers occurred. If E 1  had been less than E 2 , then E 2  remains unchanged. The acquisition search continues with epoch- 3  and so on, until all epoch values have been tried. In the end, E 2  will represent the largest correlation value found, and the sequential mode logic will know which epoch value to use. 
     There is the possibility that proper alignment was not achieved because a correlation value mistake was made due to noise. To avoid this, the entire acquisition process is repeated two more times. PN synchronization success is declared if the epoch chosen is the same for all three tries (or two out of three tries). 
     Once PN 1  has been synchronized with the received signal, the REMOTE is ready to detect data. Remember, throughout the acquisition period the LOCAL interrogator has been transmitting with d(t) equal to 1. This period will last for several hundred data bit periods. The start of a message packet will be indicated by a data transition from 1 to −1, followed by a block of unique bits which the data management logic will recognize as packet synchronization. 
     Data detection itself takes place as the received signal  1 +d(t)PN 1 (t) is gated  30  by the tag&#39;s generated PN 1  and sent to LPF 2   35 . Likewise, it is gated by PN 1 &#39;s logical compliment  37  and sent to LPF 1   38 . When d(t) is equal to 1, the input to LPF 1  is the waveform seen in FIG. 13, and the corresponding input to LPF 2  is seen as FIG.  14 . The average value of the waveform sent to LPF 2  is clearly greater. When d(t) is equal to −1, the inputs to LPF 1  and LPF 2  are as shown in FIG.  15  and FIG. 16, respectively. Here, the average value of the waveform sent to LPF 1  is greater. 
     The two low pass filters, LPF 1  and LPF 2 , are approximations to matched filters (if a simple RC filter is used, then RC=1.3/(2pi*Td)). They function to maximize the SNR at the end of the bit period, Td, for input to the comparator  39 . End of bit period timing is provided by a state AND gate  40  operating from the PN 1  generator. This gate toggles a J-K flip-flop  41  which temporarily stores the value of d(t) until it can be transferred into data management logic  42  and RAM  43 . 
     Each sequential data bit provides its own detection reference to the comparator. Also, as the received signal level varies due to movement of the tag or interrogators or changes in the RF environment, the data detection decision reference is self-adjusting, giving rise to the designation “per-bit” reference for data detection. 
     Once the forward link data has been detected, the data management logic  42  executes required actions and assembles the return link data packet message. 
     The transmitter within the tag will generate the return link carrier by multiplying the PN 1  clock frequency produced by the locked multivibrator  27  by a factor M using a frequency multiplying circuit  44 . This carrier is then amplified  45  and pulse amplitude modulated  46  by the return data packet waveform. Although an amplitude modulation scheme is shown, other modulation schemes using spread spectrum techniques are possible and desirable. This signal is then sent to the antenna  47 . The power amplifier stage may be less than 0.5 milliwatts due to the power available within the tag. Because of this low radiated power, spectrally spreading the return link carrier may not be necessary, and an additional pseudo-noise waveform analogous to PN 1  need not further encrypt d(t). When higher powers are desired, a spectrally spread scheme would be used to satisfy FCC unlicensed rules. To extend the range of the system, spread spectrum techniques are desirable in the tag transmitter. 
     While the preferred embodiments of the invention have been described, modifications can be made and other embodiments may be devised without departing from the spirit of the invention and the scope of the appended claims.