Abstract:
According to some embodiments, an electronic drive circuit is disclosed. The electronic drive circuit includes an energy storage device and a first bridge circuit coupled to the energy storage device. The first bridge circuit includes at least one leg having two switches. The electronic drive circuit also includes a transformer. The transformer includes a first winding coupled to the first bridge circuit and a second winding coupled to the energy storage device through a center tap. The electronic drive circuit further includes a second bridge circuit coupled to the second winding of the transformer. The second bridge circuit includes a pair of switches operable to conduct in both directions and block voltage in both directions. The electronic drive circuit additionally includes a DC bus coupled to the second bridge circuit and a controller, which is configured to buck or boost a DC voltage from the energy storage device to supply to the DC bus as well as buck or boost a DC voltage from the DC bus to supply to the energy storage device.

Description:
BACKGROUND OF THE INVENTION 
       [0001]    Embodiments of the present invention relate generally to electric drive systems, including hybrid and electric vehicles, and to stationary drives that are subject to transient or pulsed loads, and more particularly, to a bi-directional buck/boost DC-DC converter for transferring energy between an electrical storage device and a DC bus of the vehicle or drive. 
         [0002]    A hybrid electric vehicle (HEV) may combine an internal combustion engine and an electric motor powered by an energy storage device, such as a traction battery, to propel the vehicle. Typically, the electric motor of an HEV is coupled between the internal combustion engine and the transmission to take advantage of the torque increase through the transmission. Such a combination may increase overall fuel efficiency by enabling the combustion engine and the electric motor to each operate in their respective ranges of increased efficiency. Electric motors, for example, may be efficient at accelerating from a standing start, while combustion engines may be efficient during sustained periods of constant engine operation, such as in highway driving. Having an electric motor to boost initial acceleration allows combustion engines in HEVs to be smaller and more fuel efficient. 
         [0003]    A purely electric vehicle (EV) typically uses stored electrical energy to power an electric motor, which propels the vehicle. EVs may use one or more sources of stored electrical energy and are configured to use energy from an external source to re-charge the traction battery or other storage devices. For example, a first source of stored energy (sometimes referred to as an “energy” source) may be used to provide longer-lasting energy while a second source of stored energy (sometimes referred to as a “power” source) may be used to provide higher-power for, for example, acceleration from standstill or boost during operation. First and second sources may include chemical-based batteries or may include ultracapacitors, as examples. Typically, the source(s) of electrical energy (energy and/or power batteries) in EVs are charged via a plug-in charger or other external energy source. With typically complete reliance on plug-in power, an EV may have increased energy storage capacity and driving range as compared to an HEV. 
         [0004]    A plug-in hybrid electric vehicle (PHEV) may include both an internal combustion engine and an electric motor powered by an energy storage device, such as a traction battery. Typically a PHEV is configured to use energy from an external source to re-charge the traction battery or other storage devices. Thus, with increased reliance on plug-in power, a PHEV may have increased energy storage capacity and driving range as compared to an HEV. 
         [0005]    There are generally two types of PHEV: parallel and series. In a parallel PHEV arrangement, the electric motor is coupled between the internal combustion engine and the transmission, enabling the combustion engine and the electric motor to each operate in respective ranges of increased efficiency, similar to an HEV. In a series PHEV arrangement, the electric motor is coupled between an energy storage device and the vehicle drive axle, while the internal combustion engine is coupled directly to the energy storage device and not to the vehicle drive axle. The series PHEV may also be referred to as an extended range electric vehicle (EREV), in reference to a purely electric drive system, having energy augmentation to the energy storage system via the internal combustion engine and via, for instance, a liquid fuel storage system. 
         [0006]    In general, EVs, HEVs, and PHEVs typically include regenerative braking to charge the energy storage devices during braking operations. Also, such vehicles may include on-road and off-road vehicles, golf carts, neighborhood electric vehicles, forklifts, and utility trucks as examples. These vehicles may use either off-board stationary battery chargers or on-board battery chargers to transfer electrical energy from a utility grid or renewable energy source to the vehicle&#39;s on-board traction battery. 
         [0007]    Such vehicles may also include DC/DC converters for stepping up (boosting) or stepping down (bucking) the voltage on the DC bus. Conventional DC/DC converters include an inductor coupled to a pair of switches and coupled to a pair of diodes. Each switch is coupled to a respective diode and each switch/diode pair forms a respective half phase module. In this topology, all of the power is processed by the converter, which leads to lower efficiency. Further, there are fewer degrees of freedom with this topology. 
         [0008]    Thus, there is a need for a highly efficient bi-directional buck/boost DC/DC converter topology which provides a wide range of output voltages for providing energy to the DC bus as well as for charging one or more energy storage devices. 
       SUMMARY OF THE INVENTION 
       [0009]    According to some embodiments, an electronic drive circuit is disclosed. The electronic drive circuit includes an energy storage device and a first bridge circuit coupled to the energy storage device. The first bridge circuit includes at least one leg having two switches. The electronic drive circuit also includes a transformer. The transformer includes a first winding coupled to the first bridge circuit and a second winding coupled to the energy storage device through a center tap. The electronic drive circuit further includes a second bridge circuit coupled to the second winding of the transformer. The second bridge circuit includes a pair of switches operable to conduct in both directions and block voltage in both directions. The electronic drive circuit additionally includes a DC bus coupled to the second bridge circuit and a controller, which is configured to buck or boost a DC voltage from the energy storage device to supply to the DC bus as well as buck or boost a DC voltage from the DC bus to supply to the energy storage device. 
         [0010]    According to some embodiments, a method for operating an electronic drive circuit is disclosed. The method includes coupling an energy storage device to a first bridge circuit. The first bridge circuit includes at least one leg having two switches. The method also includes coupling a first winding of a transformer to the first bridge circuit and coupling a second winding of the transformer to the energy storage device through a center tap. The method further includes coupling a second bridge circuit to the second winding of the transformer. The second bridge circuit includes a pair of switches operable to conduct in both directions and block voltage in both directions. The method additionally includes coupling a DC bus to the second bridge circuit, and configuring a controller to buck or boost a DC voltage output from the energy storage device to supply to the DC bus as well as buck or boost a DC voltage output from the DC bus to supply to the energy storage device. 
         [0011]    According to some embodiments, an electric vehicle is disclosed. The vehicle includes an energy storage device and a first bridge circuit coupled to the energy storage device. The first bridge circuit includes at least one leg having two switches. The vehicle also includes a resonant circuit coupled to the first bridge circuit and a transformer, where the transformer includes a first winding coupled to the resonant circuit and a second winding coupled to the energy storage device through a center tap. The vehicle further includes a second bridge circuit coupled to the second winding of the transformer. The second bridge circuit includes a pair of switches operable to conduct in both directions and block voltage in both directions. The vehicle additionally includes a DC bus coupled to the second bridge circuit and a traction drive of the vehicle coupled to the DC bus. The vehicle also includes a controller configured to buck or boost a DC voltage from the energy storage device to supply to the DC bus as well as buck or boost a DC voltage from the DC bus to supply to the energy storage device. 
         [0012]    Various other features and advantages will be made apparent from the following detailed description and the drawings. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0013]    In order for the advantages of the invention to be readily understood, a more particular description of the invention briefly described above will be rendered by reference to specific embodiments that are illustrated in the appended drawings. Understanding that these drawings depict only exemplary embodiments of the invention and are not, therefore, to be considered to be limiting its scope, the invention will be described and explained with additional specificity and detail through the use of the accompanying drawings, in which: 
           [0014]      FIG. 1  is a schematic diagram of a traction system according to an embodiment of the present invention; 
           [0015]      FIG. 2  is a pulse sequence diagram illustrating a gating sequence for discharging an energy storage device by stepping up a voltage of that energy storage device according to an embodiment of the present invention; 
           [0016]      FIG. 3  is a pulse sequence diagram illustrating a gating sequence for discharging an energy storage device by stepping down the voltage of that energy storage device according to an embodiment of the present invention; 
           [0017]      FIG. 4  is a pulse sequence diagram illustrating a gating sequence for charging an energy storage device by stepping up a DC bus voltage according to an embodiment of the present invention; 
           [0018]      FIG. 5  is a pulse sequence diagram illustrating a gating sequence for charging an energy storage device by stepping down the DC bus voltage according to an embodiment of the present invention; 
           [0019]      FIG. 6  is a schematic diagram of a traction system according to another embodiment of the present invention; and 
           [0020]      FIG. 7  is a schematic diagram of a traction system according to another embodiment of the present invention. 
       
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       [0021]    Disclosed herein is a partial power processing bi-directional, buck/boost converter topology. Bi-directional power flow allows battery charging during regenerative braking mode. The converter may be operated in either a step up (boost) or step down (buck) mode, thus allowing optimization of a DC bus voltage according to required motor speed. 
         [0022]      FIG. 1  illustrates a schematic diagram of a traction system  100  according to an embodiment of the present invention. The traction system  100  may be included in a vehicle such as an electric vehicle (EV), hybrid electric vehicle (HEV), or plug-in hybrid electric vehicle (PHEV). The traction system  100  may alternatively be included in a stationary electric drive system. 
         [0023]    Traction system  100  includes an energy storage device  102 . As nonlimiting examples, energy storage device  102  may be a battery, a fuel cell, or an ultracapacitor. 
         [0024]    The energy storage device  102  is coupled via a DC link  104  to a first bridge circuit  106 . The first bridge circuit  106  includes four switches  108 - 114 . As nonlimiting examples, switches  108 - 114  may be Si or SiC MOSFETs, IGBTs, MCTs, Thyristors, GTOs, IGCTs, cascode switches with SiC JFETs and Si MOSFETs or with GaN HEMTs and Si switches, or cross switches such as a hybrid Si/SiC device: Si IGBTs and SiC Schottky diodes. The first bridge circuit  106  further includes four diodes  116 - 122  each coupled in parallel with a corresponding switch  108 - 114 . The first bridge circuit  106  is coupled to a primary winding  124  of a transformer  126  via junctions  128  and  130 . 
         [0025]    In an exemplary embodiment, the first bridge circuit  106  may be used to pulse width modulate (PWM) an input voltage from the energy storage device  102  for input into the primary winding  124  of the transformer  126 . This PWM function of the first bridge circuit  106  is controlled by a controller  132 , which is coupled to switches  108 - 114  via control lines  134 . The controller  132  controls the action of switches  108 - 114  by controlling a duty cycle of the PWM (i.e., the length of time the switches stay on/off). In this way, the input voltage from the energy storage device  102  is converted into a PWM voltage input into the primary winding  124  for conversion by the transformer  126  into a secondary voltage in the secondary winding  136  of the transformer  126 . In alternative embodiments, the first bridge circuit  106  may be operated by variable frequency or phase shift control. Any of these control methods may be used to regulate the output voltage to a set reference value. Variable frequency control may be used to achieve higher operating efficiency over a wide operating range, while PWM or phase shift may be used closer to the limits of the operating range. 
         [0026]    The first bridge circuit  106  may also operate as a rectifier to output a voltage for charging the energy storage device  102 . In this case, switches  108 - 114  remain off through controller  132 . Alternatively, switches  108 - 114  may be switched at a specified phase shift angle to regulate power flow to charge the battery. These operations would occur during regenerative braking, as a nonlimiting example. 
         [0027]    The secondary winding  136  of transformer  126  is coupled to a second bridge circuit  138 . The secondary voltage generated by the secondary winding  136  of the transformer  126  is input into the second bridge circuit  138 . The input voltage of the energy storage device  102  is also input into a center tap  139  of the secondary winding  136  of the transformer  126  to allow some input power to bypass the first bridge circuit  106 , thereby increasing the overall efficiency of the traction system  100 . 
         [0028]    Partial power processing means the converter rating may be made lower than the full power rating. Depending on the battery voltage range and maximum voltage required for the DC bus, the converter may be designed at about 67% the full rating as an example. Partial power processing leads to higher efficiency as part of the input power is fed directly to the output at 100% efficiency. 
         [0029]    The second bridge circuit  138  includes four switches  140 - 146 . As nonlimiting examples, switches  140 - 146  may be Si or SiC MOSFETs, IGBTs, MCTs, Thyristors, GTOs, IGCTs, cascode switches with SiC JFETs and Si MOSFETs or with GaN HEMTs and Si switches, or cross switches such as a hybrid Si/SiC device: Si IGBTs and SiC Schottky diodes. The switches  140 - 146  may alternatively be reverse blocking or reverse conducting IGBTs. The second bridge circuit  138  further includes four diodes  148 - 154  each coupled in parallel with a corresponding switch  140 - 146 . The switches  140 - 146  are AC switches which may conduct in both directions and block voltage in both directions depending on the operation of the switches  140 - 146 . Controller  132  is further coupled to switches  140 - 146  via control lines  156  to control the action of switches  140 - 146  to operate the second bridge circuit  138  as a rectifier. An inductor  158  and capacitor  160  are also coupled to the second bridge circuit  138  to filter the output. In this manner, the second bridge circuit  138  outputs a load voltage to a DC bus  162 . 
         [0030]    In an exemplary embodiment, the DC bus  162  is coupled to a traction drive  164 . Traction drive may include an inverter  166  coupled to a traction motor  168 . However, in alternative embodiments (not shown), the DC bus  162  may be coupled to a second energy storage device, an electric drive including an inverter and electric motor, or another DC/DC converter to further convert the DC voltage for DC loads such as stepper motors or auxiliary loads (e.g. air conditioning, power windows, or a stereo system). 
         [0031]    The second bridge circuit  138  may also be used to PWM a voltage from the DC bus  162  to the secondary winding  136  of transformer  126 . The controller  132  controls the action of switches  148 - 154  by controlling a PWM duty cycle. In this way, the voltage from the DC bus  162  is converted into a PWM voltage input into the secondary winding  136  for conversion by the transformer  126  into another voltage in the primary winding  124  of the transformer  126  for charging the energy storage device  102 . In alternative embodiments, the second bridge circuit  138  may be operated by variable frequency or phase shift control. Any of these control methods may be used to regulate the output voltage to a set reference value. Variable frequency control may be used to achieve higher operating efficiency over a wide operating range, while PWM or phase shift may be used closer to the limits of the operating range. PWM, variable frequency, or phase shift control would occur during regenerative braking, as a nonlimiting example. 
         [0032]    Traction system  100  may further include a resonant circuit  170  between the first bridge circuit  106  and primary winding  124  of the transformer  126 . In the exemplary embodiment shown in  FIG. 1 , resonant circuit  170  is an LLC circuit, including two inductors (LL)  172  and  174 , and capacitor (C)  176 . In alternative embodiments, the resonant circuit  170  may be an LC-parallel circuit or an LCC circuit. Including the resonant circuit  170  in traction system  100  provides more degrees of freedom for controlling power flow and voltage regulation. Resonant circuit  170  also provides soft switching, improving overall efficiency. Switching frequency of switches  108 - 114  dictates the gain of a resonant tank. This can provide step up or step down operation depending on whether the switching frequency is below the series resonant frequency (step-up) or above the series resonant frequency (step-down). The voltage gain of the resonant converter is dependent on the operating frequency. For the LLC resonant circuit as a non-limiting example, the voltage gain is greater than 1 below the series resonant frequency which provides step up capability. The voltage gain is less than 1 above the series resonant frequency which provides step down operation. In PWM control, a duty ratio of 0.5 provides the highest gain, while any other duty ratio gives lower gain. When both step up and step down capability are combined, there is more flexibility in regulating the terminal voltage as well as improving efficiency by achieving zero voltage switching transitions for the active devices. 
         [0033]      FIG. 2  is a pulse sequence diagram  178  illustrating a gating sequence in a step up mode during discharge according to an embodiment of the present invention. In the step up mode, power from the converter is added to the output to increase the voltage. The pulse sequence diagram  166  shows the various waveforms during a single period T. 
         [0034]    In a step up mode during discharge of the energy storage device  102 , the first bridge circuit  106  may operate as a PWM converter and the second bridge circuit  138  operates as a rectifier. As such, switches  108 - 114  are pulse width modulated as shown in  FIG. 2 , and switches  140 - 142  are turned on while switches  144 - 146  are turned off. When the switches  108 - 114  of first bridge circuit  106  are conducting, a voltage is impressed on the secondary winding  136  of the transformer  126 . The instantaneous output at the diodes  152 - 154  is the input voltage of the energy storage device  102  plus the transformer secondary voltage, which is equal to the input voltage divided by the transformer turns ratio N p /N s . By varying the PWM duty cycle D from 0% to 100%, the average voltage output can be controlled between a minimum of the input voltage and a maximum of the input voltage plus the boost provided by the secondary voltage output of the transformer  126 . The maximum gain occurs at 50% duty cycle and symmetrically drops for higher or lower values of the duty cycle. The inductor  158  and capacitor  160  operate as filters to smooth the output voltage. The output voltage thus will be V out =V in +(Gain)*V in . For PWM operation, the voltage gain (Gain) is equal to 2*D*(N s /N p ). 
         [0035]    The first bridge circuit may also operate as a variable frequency converter. For variable frequency operation when utilizing an LLC resonant converter, the Gain is equal to: 
         [0000]    
       
         
           
             Gain 
             = 
             
               
                 ( 
                 
                   
                     N 
                     s 
                   
                   
                     N 
                     p 
                   
                 
                 ) 
               
               * 
               
                 1 
                 
                   
                     
                       
                         ( 
                         
                           1 
                           + 
                           
                             1 
                             λ 
                           
                           - 
                           
                             1 
                             
                               λ 
                                
                               
                                   
                               
                                
                               
                                 f 
                                 n 
                                 2 
                               
                             
                           
                         
                         ) 
                       
                       2 
                     
                     + 
                     
                       
                         
                           Q 
                           2 
                         
                          
                         
                           ( 
                           
                             
                               f 
                               n 
                             
                             - 
                             
                               1 
                               
                                 f 
                                 n 
                               
                             
                           
                           ) 
                         
                       
                       2 
                     
                   
                 
               
             
           
         
       
     
         [0000]    and V out =V in (1±Gain). In this calculation, f n  is the normalized switching frequency 
         [0000]    
       
         
           
             
               ( 
               
                 
                   f 
                   n 
                 
                 = 
                 
                   
                     f 
                     s 
                   
                   
                     f 
                     r 
                   
                 
               
               ) 
             
             , 
           
         
       
     
         [0000]    λ is the inductance ratio, and Q is the quality factor. If first bridge circuit operates with a combination of PWM and frequency control, the voltage gain will be a combined value. If first bridge circuit operates with a combination of PWM and frequency control, the voltage gain will be a combined value. 
         [0036]    For an LLC resonant converter topology, in order to achieve zero voltage switching (ZVS), the switching frequency need to be higher than the resonant frequency, such that the resonant tank current lags the resonant tank voltage. A lagging tank current leads to a negative device current before the device is turned on. This negative current discharges the device/snubber capacitance and then flows through the anti-parallel diode. Therefore, the device voltage drops to zero before it is turned on and thus ZVS is achieved. 
         [0037]    The device turn off current is limited to the magnetizing current if the switching frequency f s  is in the range f m &lt;f s &lt;f r . Here, f m  is the overall resonant frequency due to the resonance between the resonant capacitor and combined inductance equal to 
         [0000]    
       
         
           
             
               f 
               m 
             
             = 
             
               1 
               
                 2 
                  
                 
                     
                 
                  
                 π 
                  
                 
                   
                     
                       ( 
                       
                         
                           L 
                           r 
                         
                         + 
                         
                           L 
                           m 
                         
                       
                       ) 
                     
                      
                     
                       C 
                       r 
                     
                   
                 
               
             
           
         
       
     
         [0000]    and f r  is the series resonant frequency due to the resonance between the inductor and capacitor in series equal to 
         [0000]    
       
         
           
             
               f 
               r 
             
             = 
             
               
                 1 
                 
                   2 
                    
                   
                       
                   
                    
                   π 
                    
                   
                     
                       
                         L 
                         r 
                       
                        
                       
                         C 
                         r 
                       
                     
                   
                 
               
               . 
             
           
         
       
     
         [0000]    Beyond f r , the turn off current becomes much higher and thus the switching loss increases at turn off. To compensate for that, snubber capacitors may be used to limit the device voltage to achieve close to zero voltage switching at turn off. 
         [0038]    The quality factor Q of the resonant circuit is 
         [0000]    
       
         
           
             
               = 
               
                 
                   z 
                   C 
                 
                 
                   R 
                   ac 
                 
               
             
             , 
           
         
       
     
         [0000]    where Z C  is the converter characteristic impedance, which is 
         [0000]    
       
         
           
             
               
                 Z 
                 C 
               
               = 
               
                 
                   
                     L 
                     r 
                   
                   
                     C 
                     r 
                   
                 
               
             
             , 
           
         
       
     
         [0000]    and R ac  is the equivalent load resistance of the series resonant converter, which is 
         [0000]    
       
         
           
             
               R 
               ac 
             
             = 
             
               
                 8 
                 
                   π 
                   2 
                 
               
                
               
                 
                   ( 
                   
                     
                       N 
                       p 
                     
                     
                       N 
                       s 
                     
                   
                   ) 
                 
                 2 
               
                
               
                 
                   R 
                   load 
                 
                 . 
               
             
           
         
       
     
         [0000]    From the gain characteristics low quality factor provides high gain selectivity below f r  and almost a flat gain characteristic beyond f r . A low quality factor also leads to low voltage across the resonant capacitor, which is reflected to the diode voltage at turn off. Changing the inductance ratio 
         [0000]    
       
         
           
             λ 
             = 
             
               
                 L 
                 m 
               
               
                 L 
                 r 
               
             
           
         
       
     
         [0000]    also has a significant effect on the converter gain characteristics. A low inductance ratio gives high frequency sensitivity but leads to more circulating current. A high inductance ratio gives low circulating current and thus better efficiency, but the gain characteristic is flat compared to switching frequency. As such, a transformer should have a low quality factor and high inductance ratio to achieve higher efficiency and low component stresses. 
         [0039]    For other types of resonant tanks, the Gain will change accordingly but the analysis is similar to the LLC resonant converter topology as described above. For an LC series resonant converter, the Gain is equal to: 
         [0000]    
       
         
           
             Gain 
             = 
             
               
                 ( 
                 
                   
                     N 
                     s 
                   
                   
                     N 
                     p 
                   
                 
                 ) 
               
               * 
               
                 
                   1 
                   
                     
                       1 
                       + 
                       
                         
                           
                             π 
                             2 
                           
                           64 
                         
                          
                         
                           
                             
                               Q 
                               2 
                             
                              
                             
                               ( 
                               
                                 
                                   f 
                                   n 
                                 
                                 - 
                                 
                                   1 
                                   
                                     f 
                                     n 
                                   
                                 
                               
                               ) 
                             
                           
                           2 
                         
                       
                     
                   
                 
                 . 
               
             
           
         
       
     
         [0000]    For an LC parallel resonant converter, the Gain is equal to: 
         [0000]    
       
         
           
             Gain 
             = 
             
               
                 ( 
                 
                   
                     N 
                     s 
                   
                   
                     N 
                     p 
                   
                 
                 ) 
               
                
               
                 
                   1 
                   
                     
                       
                         
                           
                             π 
                             2 
                           
                           64 
                         
                          
                         
                           
                             ( 
                             
                               1 
                               - 
                               
                                 f 
                                 n 
                                 2 
                               
                             
                             ) 
                           
                           2 
                         
                       
                       + 
                       
                         
                           f 
                           n 
                           2 
                         
                          
                         
                           ( 
                           
                             1 
                             
                               Q 
                               2 
                             
                           
                           ) 
                         
                       
                     
                   
                 
                 . 
               
             
           
         
       
     
         [0000]    For an LCC series-parallel resonant converter, the Gain is equal to: 
         [0000]    
       
         
           
             Gain 
             = 
             
               
                 ( 
                 
                   
                     N 
                     s 
                   
                   
                     N 
                     p 
                   
                 
                 ) 
               
                
               
                 
                   1 
                   
                     
                       
                         
                           
                             π 
                             2 
                           
                           64 
                         
                          
                         
                           
                             ( 
                             
                               1 
                               + 
                               
                                 
                                   
                                     C 
                                     P 
                                   
                                   
                                     C 
                                     S 
                                   
                                 
                                  
                                 
                                   ( 
                                   
                                     1 
                                     - 
                                     
                                       f 
                                       n 
                                       2 
                                     
                                   
                                   ) 
                                 
                               
                             
                             ) 
                           
                           2 
                         
                       
                       + 
                       
                         
                           
                             Q 
                             2 
                           
                            
                           
                             ( 
                             
                               
                                 f 
                                 n 
                               
                               - 
                               
                                 1 
                                 
                                   f 
                                   n 
                                 
                               
                             
                             ) 
                           
                         
                         2 
                       
                     
                   
                 
                 . 
               
             
           
         
       
     
         [0040]      FIG. 3  is a pulse sequence diagram  180  illustrating a gating sequence in a step down mode during discharge according to an embodiment of the present invention. In the step down mode, the input voltage is greater than the desired output voltage. The pulse sequence diagram  168  shows the various waveforms during a single period T. 
         [0041]    In a step down mode during discharge of the energy storage device  102 , the first bridge circuit  106  operates as a full wave diode rectifier and the second bridge circuit  138  operates as a PWM converter. As such, switches  108 - 114  and  144 - 146  are turned off, while switches  140 - 142  are pulse width modulated as shown in  FIG. 3 . In step down operation, the secondary winding  136  of the transformer  126  acts as a primary winding, whereas the primary winding  124  of the transformer  126  acts as a secondary winding. 
         [0042]    The output voltage of the second bridge circuit  138  is recirculated back to the first bridge circuit  106 . Varying the duty cycle of the switches  140 - 142  controls the ratio between the input and output voltage of the second bridge circuit  138 . Since the input voltage of the energy storage device  102  fixes the output voltage of the second bridge circuit  138 , the effect of varying the duty cycle is to vary the voltage drop between the first bridge circuit  106  and the output voltage. Switches  140 - 142  are controlled such that either one or both switches are always turned on; they are never both off simultaneously, even during the switching cycle. When both switches are conducting there is no voltage drop across the input of the second bridge circuit  138 , so the output voltage is equal to its input voltage. When one switch is open, the voltage across the transformer  126  will be equal to the input voltage of the energy storage device  102  since it is clamped to that value by the first bridge circuit  106 . Therefore, the input voltage across the second bridge circuit  138  will be equal to the input voltage of the energy storage device  102  divided by the transformer turns ratio. Varying the duty cycle controls the average voltage dropped across the second bridge circuit  138 , and therefore the output voltage. The output voltage thus will be V out =V in −2*V in *(1−D)*(N s /N p ). 
         [0043]    The second bridge circuit may also operate as a variable frequency converter. For variable frequency operation when utilizing a LLC resonant converter, the Gain is equal to: 
         [0000]    
       
         
           
             Gain 
             = 
             
               
                 ( 
                 
                   
                     N 
                     s 
                   
                   
                     N 
                     p 
                   
                 
                 ) 
               
               * 
               
                 1 
                 
                   
                     
                       
                         ( 
                         
                           1 
                           + 
                           
                             1 
                             
                               λ 
                                
                               
                                   
                               
                             
                           
                           - 
                           
                             1 
                             
                               λ 
                                
                               
                                   
                               
                                
                               
                                 f 
                                 n 
                                 2 
                               
                             
                           
                         
                         ) 
                       
                       2 
                     
                     + 
                     
                       
                         
                           Q 
                           2 
                         
                          
                         
                           ( 
                           
                             
                               f 
                               n 
                             
                             - 
                             
                               1 
                               
                                 f 
                                 n 
                               
                             
                           
                           ) 
                         
                       
                       2 
                     
                   
                 
               
             
           
         
       
     
         [0000]    and V out =V in (1±Gain). 
         [0044]      FIG. 4  is a pulse sequence diagram  182  illustrating a gating sequence for charging the energy storage device  102  by stepping up an input voltage from the DC bus  162  according to an embodiment of the present invention. In this case, the first bridge circuit  106  operates as a full wave rectifier and the second bridge circuit  138  operates as a PWM converter. As such, switches  108 - 114  are turned off, switches  140 - 142  are turned on, and switches  144 - 146  are pulse width modulated as shown in  FIG. 4 . 
         [0045]    The output voltage of the first bridge circuit  106  is recirculated back to the second bridge circuit  138 . Varying the duty cycle of the switches  144 - 146  controls the ratio between the input and output voltage of the first bridge circuit  106 . Since the input voltage of the DC bus  162  fixes the output voltage of the first bridge circuit  106 , the effect of varying the duty cycle is to vary the voltage drop between the second bridge circuit  138  and the output voltage. Switches  144 - 146  are controlled such that either one or both switches are always turned on; they are never both off simultaneously, even during the switching cycle. When both switches are conducting there is no voltage drop across the input of the second bridge circuit  138 , so the output voltage is equal to its input voltage. When one switch is open, the voltage across the transformer  126  will be equal to the input voltage of the DC bus  162  since it is clamped to that value by the second bridge circuit  138 . Therefore, the input voltage across the first bridge circuit  106  will be equal to the input voltage of the DC bus  162  divided by the transformer turns ratio. By varying the PWM duty cycle, the average voltage output to charge the energy storage device  102  can be controlled between a minimum of the input voltage and a maximum of the input voltage plus the boost provided by the voltage output from the transformer  126 . The output voltage thus will be V out =V in +2*D*(N s /N p )*V in . 
         [0046]    The second bridge circuit may also operate as a variable frequency converter. For variable frequency operation when utilizing an LLC resonant converter, the Gain is equal to: 
         [0000]    
       
         
           
             Gain 
             = 
             
               
                 ( 
                 
                   
                     N 
                     s 
                   
                   
                     N 
                     p 
                   
                 
                 ) 
               
               * 
               
                 1 
                 
                   
                     
                       
                         ( 
                         
                           1 
                           + 
                           
                             1 
                             
                               λ 
                                
                               
                                   
                               
                             
                           
                           - 
                           
                             1 
                             
                               λ 
                                
                               
                                   
                               
                                
                               
                                 f 
                                 n 
                                 2 
                               
                             
                           
                         
                         ) 
                       
                       2 
                     
                     + 
                     
                       
                         
                           Q 
                           2 
                         
                          
                         
                           ( 
                           
                             
                               f 
                               n 
                             
                             - 
                             
                               1 
                               
                                 f 
                                 n 
                               
                             
                           
                           ) 
                         
                       
                       2 
                     
                   
                 
               
             
           
         
       
     
         [0000]    and V out =V in (1±Gain). If first bridge circuit operates with a combination of PWM and frequency control, the voltage gain will be a combined value. 
         [0047]      FIG. 5  is a pulse sequence diagram  184  illustrating a gating sequence for charging an energy storage device by stepping down the input voltage according to an embodiment of the present invention. In this case, both bridge circuits  106  and  138  are operating as variable frequency converters. As such, switches  140 - 142  are turned off, and switches  108 - 114  and  144 - 146  are phase shifted as shown in  FIG. 4 . Variable frequency is used to adjust the gain between the two bridges  106  and  138 . The phase shift between the two bridges  106  and  138  regulates the amount of current flowing between the two sides. This mode of operation is similar to a resonant dual active bridge. Switches  140 - 142  are turned off in order to maintain power flow direction to charge the energy storage device  102 . Soft switching still occurs, where the first bridge circuit  106  operates under zero current switching while the second bridge circuit  138  operates with zero voltage switching. The Gain is equal to: 
         [0000]    
       
         
           
             Gain 
             = 
             
               
                 ( 
                 
                   
                     N 
                     s 
                   
                   
                     N 
                     p 
                   
                 
                 ) 
               
               * 
               
                 1 
                 
                   
                     
                       
                         ( 
                         
                           1 
                           + 
                           
                             1 
                             
                               λ 
                                
                               
                                   
                               
                             
                           
                           - 
                           
                             1 
                             
                               λ 
                                
                               
                                   
                               
                                
                               
                                 f 
                                 n 
                                 2 
                               
                             
                           
                         
                         ) 
                       
                       2 
                     
                     + 
                     
                       
                         
                           Q 
                           2 
                         
                          
                         
                           ( 
                           
                             
                               f 
                               n 
                             
                             - 
                             
                               1 
                               
                                 f 
                                 n 
                               
                             
                           
                           ) 
                         
                       
                       2 
                     
                   
                 
               
             
           
         
       
     
         [0000]    and V out =V in (1±Gain). 
         [0048]      FIG. 6  is a schematic diagram of a traction system  186  according to an alternative embodiment of the present invention. Elements and components common to traction system  100  are referred to herein with similar part numbering as appropriate. In this half bridge embodiment, switches  108 - 110  are replaced with capacitors  188 - 190  in a first half bridge  192 . The secondary winding  136  of transformer  126  is coupled to a second half bridge  194  via junctions  196  and  198 . The second half bridge  194  includes four switches  200 - 206  coupled via junction  196 , each switch coupled in parallel with a corresponding diode  208 - 214 . The second half bridge  194  further includes two capacitors  216 - 218  coupled via junction  198 . A resonant circuit  170  may be included between the first half bridge  192  and the primary winding  124  of the transformer  126  as well. 
         [0049]    Traction system  186  operates similar to the embodiments discussed previously. For discharge step up mode corresponding to  FIG. 2 , switches  112  and  114  will be switching (via either PWM or variable frequency control) on the first bridge  192 . On the second bridge  194 , switches  200  and  204  are turned off while switches  202  and  206  are turned on. For discharge step down mode corresponding to  FIG. 3 , switches  112 - 114 ,  200 , and  204  are turned off, while switches  202  and  206  are switching (via either PWM or variable frequency control). For charge step up mode corresponding to  FIG. 4 , switches  112 - 114 ,  202 , and  206  are turned off, while switches  200  and  204  are switching (via either PWM or variable frequency control to regulate charging current and gain). For charge step down mode corresponding to  FIG. 5 , switches  202  and  206  are turned off, while switches  112 - 114 ,  200 , and  204  are switching (via either PWM or variable frequency control). The variable frequency control regulates the gain, and the phase shift between first and second bridges  192  and  194  regulates the charging current. In this embodiment, the second bridge  194  operates as a voltage doubler rectifier, and as such, can be used for applications where higher voltage gain is needed. 
         [0050]      FIG. 7  is a schematic diagram of a traction system  222  according to another alternative embodiment of the present invention. Elements and components common to traction system  100  are referred to herein with similar part numbering as appropriate. In this full bridge embodiment, an additional capacitor  224  is included in a first full bridge  226 . The secondary winding  136  of transformer  126  is coupled to a second full bridge  228  via junctions  230  and  232 . The second full bridge  228  includes four switches  234 - 240  coupled via junction  230 , each switch coupled in parallel with a corresponding diode  242 - 248 . The second full bridge  228  further includes four switches  250 - 256  coupled via junction  232 , each switch coupled in parallel with a corresponding diode  258 - 264 . A resonant circuit  170  may be included between the first full bridge  226  and the primary winding  124  of the transformer  126  as well. 
         [0051]    Traction system  222  operates similar to the embodiments discussed previously, but is better suited for higher power applications. For discharge step up mode corresponding to  FIG. 2 , switches  108 - 114  are switching (via PWM or variable frequency control). The second bridge  228  operates in a rectifier mode, such that switches  234 ,  238 ,  250 , and  254  are turned off while switches  236 ,  240 ,  252 , and  256  are turned on. For discharge step down mode corresponding to  FIG. 3 , switches  108 - 114  are turned off. Switches  234 ,  238 ,  250  and  254  are also turned off, while switches  236 ,  240 ,  252 , and  256  are switching (via PWM or variable frequency control to regulate voltage gain). For charge step up mode corresponding to  FIG. 4 , switches  108 - 114  are turned off. Switches  236 ,  240 ,  252 , and  256  are turned off while switches  234 ,  238 ,  250 , and  254  are switching (via PWM or variable frequency control to regulate voltage gain and charging current). For charge step down mode corresponding to  FIG. 5 , switches  108 - 114  are switching (via PWM and/or variable frequency control). Switches  236 ,  240 ,  252 , and  256  are turned off while switches  234 ,  238 ,  250 , and  254  are switching (via PWM and/or variable frequency control to regulate the voltage gain). Charging current is controlled by the phase shift between switches  234 ,  238 ,  250 , and  254  and the first bridge switches  108 - 114 . 
         [0052]    It is understood that the above-described embodiments are only illustrative of the application of the principles of the present invention. The present invention may be embodied in other specific forms without departing from its spirit or essential characteristics. All changes that come within the meaning and range of equivalency of the claims are to be embraced within their scope. Thus, while the present invention has been fully described above with particularity and detail in connection with what is presently deemed to be the most practical and preferred embodiment of the invention, it will be apparent to those of ordinary skill in the art that numerous modifications may be made without departing from the principles and concepts of the invention as set forth in the claims.