Abstract:
A method and apparatus for generating a variable output voltage from a voltage reference circuit is disclosed. A voltage reference circuit includes a first voltage generator configured for generating a first voltage signal having a negative temperature coefficient and a second voltage generator configured for generating a second voltage signal having a positive temperature coefficient. The voltage reference circuit further includes a current generator configured for supplying a reference current to the first voltage generator and the second voltage generator. A comparator configured for comparing the first voltage signal to the second voltage signal generates a comparison result to modify the reference current with a current change related to the result of the comparison. Finally, the voltage reference circuit also includes an output terminal operably coupled to the current generator, wherein the output terminal comprises a voltage that is a voltage differential above a bandgap voltage and substantially independent of temperature change.

Description:
BACKGROUND OF THE INVENTION  
     Field of the Invention  
       [0001]     The present invention relates to voltage reference circuits. More specifically, the present invention relates to circuits and methods for generating a variable reference voltage from a bandgap reference.  
         [0002]     Many systems that manipulate and generate analog and digital signals need precise, stable voltage and current references defining bias points for these systems. In many cases, these voltage references must be in addition to and independent from a supply voltage for the circuit. In Dynamic Random Access Memories (DRAM), as well as other semiconductor devices, some of these applications are in areas such as sense amplifiers, input signal level sensors, phase locked loops, delay locked loops, and various other analog circuits.  
         [0003]     Many techniques exist for generating these voltage references. Traditional bias generation techniques vary from a simple resistor voltage divider, to the voltage drop generated by forward biased diodes, to reverse-biased Zener diodes, to more precise bandgap reference circuits. These reference voltages may typically need to be independent from a source supply voltage and relatively constant across temperature variations.  
         [0004]     A voltage reference may be created from a traditional and simple voltage divider circuit using resistors in series. Unfortunately, the resultant reference voltage is a function of the supply voltage and controlling the precision of the resistors may be difficult. Voltage dividers are, therefore, not an adequate solution when supply independence is required.  
         [0005]     The voltage drop across a diode may be used to generate a voltage supply independent reference voltage. However, a diode voltage drop is temperature dependent and inadequate for systems where the reference voltage must be substantially constant over a wide range of temperatures.  
         [0006]     Complementary MOS (CMOS) circuits are often used to generate supply independent reference voltages using transistor threshold voltages (Vt) to generate a reference. These circuits typically have the advantage of being small in area, relatively simple, and relatively independent from the supply voltage. However, as with diode references, Vt referenced bias sources typically vary with changes in temperature.  
         [0007]     Bandgap reference sources are quite flexible and may generate reference voltages that are substantially voltage supply independent and substantially temperature independent. However, conventional bandgap reference circuits generate a voltage at the bandgap of silicon, or integer multiples of the bandgap voltage.  
         [0008]     A circuit diagram of a conventional bandgap reference  10  is shown in  FIG. 1 . The bandgap reference includes a p-channel transistor  12  configured as a current source, an amplifier  15 , two diode connected bipolar transistors ( 28  and  38 ), and resistors ( 22 ,  32 , and  36 ). The bipolar transistors ( 28  and  38 ) are configured with junction areas of relative size such that the first bipolar transistor  28  has a P—N junction area with a relative size of one, and the second bipolar transistor  38  has a P—N junction area that is N times the size of bipolar transistor  28 .  
         [0009]     Generally, a bandgap reference is derived from the principal that two diodes of different sizes, but with the same emitter current, will have different current densities and, as a result, slightly different voltage drops across the P—N junction. Furthermore, P—N junctions have a negative temperature coefficient wherein changes in the voltage drop across the P—N junction are inversely proportional to changes in temperature. In other words, as temperature rises, the voltage drop across a P—N junction falls. For example, for silicon, the voltage drop across a P—N junction is inversely proportional to temperature changes at about −2.2 mV/° C.  
         [0010]     In operation, the feedback on the amplifier  15  operates to develop a steady state wherein the inverting input node  20  and the non-inverting input node  30  are maintained at substantially the same voltage potential. If the inputs are not at the same potential, the amplifier  15  acts to reduce or increase the voltage on a feedback node  18 . In turn, the voltage on the feedback node  18  will increase or decrease the current through the p-channel transistor  12 . Thus, for a circuit wherein resistors  22  and  32  have the same value, the voltage drop across the first bipolar transistor  28  is equal to the combination of the voltage drop across the second bipolar transistor  38  and the voltage drop across resistor  36 . As a result, the voltage drop across resistor  36  represents the difference between the voltage drop across the first transistor  28  and the voltage drop across the second transistor  38 . This difference generally may be referred to as ΔV be  indicating that it represents the difference in voltage drop between the two bipolar transistors  28  and  38 . ΔV be  may also be referred to as a voltage that is Proportional to Absolute Temperature (PTAT) because the voltage adjusts in proportion to temperature change with a positive temperature coefficient substantially opposite to the negative temperature coefficient of the first bipolar transistor  28  such that the output signal  40  remains substantially temperature independent.  
         [0011]     Due to the negative temperature coefficient for diodes, as temperature rises, the V be  of the first bipolar transistor  28  decreases at a higher rate than the V be  decrease of the second bipolar transistor  38 . Consequently, to keep the feedback loop in a steady state, the ΔV be  across resistor  36 , has a direct temperature correlation (i.e., voltage change increases as temperature increases). When in the steady state, the circuit generates a resulting output signal  40  substantially equal to the bandgap voltage of silicon, which is about 1.25 volts.  
         [0012]      FIG. 2  illustrates the negative temperature coefficient  25  of the first bipolar transistor  28 , the positive temperature coefficient  35  resulting from the V PTAT , and the substantially temperature independent output voltage  45 .  
         [0013]     A circuit diagram of another conventional bandgap reference  60  is shown in  FIG. 3 . The bandgap reference  60  of  FIG. 3  may be configured to generate a reference voltage that is twice the bandgap voltage. The bandgap reference  60  includes a p-channel transistor  62  configured as a current source, an amplifier  65 , four diode connected bipolar transistors ( 78 ,  79 ,  88 , and  89 ), and resistors ( 72 ,  82 , and  86 ). In this circuit, bipolar transistor  78  and bipolar transistor  79  are connected in series to create two diode voltage drops. Similarly, bipolar transistor  88  and bipolar transistor  89  are connected in series to create two diode voltage drops. The area of the P—N junctions of bipolar transistors  88  and  89  are larger, such as, for example, N times as large as the area of the P—N junctions of bipolar transistors  78  and  79 .  
         [0014]     The feedback for the circuit of  FIG. 3  operates similar to that of the circuit of  FIG. 1 . As a result, the voltage drop across resistor  86  is set to be about the difference between the two-diode voltage drop across bipolar transistors  78 / 79  and the two-diode voltage drop across bipolar transistors  88 / 89 . The resulting voltage on the output signal  90  is about twice the silicon bandgap voltage (i.e., about 2.5 volts). This circuit may be expanded by using more than two transistors in series to create voltage references that are integer multiples of the bandgap voltage.  
         [0015]     However, there is a need for a reference voltage generator that is substantially temperature independent, substantially supply voltage independent, and that may generate a variable output above the bandgap voltage that is not an integer multiple of the silicon bandgap voltage.  
       BRIEF SUMMARY OF THE INVENTION  
       [0016]     The present invention in a number of embodiments includes methods and apparatuses for generating a reference voltage that is substantially temperature independent, substantially supply voltage independent, and at a voltage output above a bandgap voltage.  
         [0017]     In one embodiment of the invention, a voltage reference circuit includes a first voltage generator configured for generating a first voltage signal having a negative temperature coefficient. The voltage reference circuit further includes a current generator configured for supplying a reference current having a positive temperature coefficient and an offset current, wherein the reference current is related to a voltage of the first voltage signal. The voltage reference circuit further includes a first resistance element operably coupled between the first voltage generator and the current generator. Finally, the voltage reference circuit includes an output signal operably coupled the current generator, wherein the output signal comprises a voltage that is a voltage offset above a bandgap voltage and substantially independent of a temperature change.  
         [0018]     In another embodiment of the invention, a voltage reference circuit comprises an amplifier having a first input, a second input, and a comparison result. The voltage reference circuit further includes a current source configured for sourcing a current related to a voltage of the comparison result, wherein an output of the current source is configured as an output signal. The voltage reference circuit further includes a first resistance element operably coupled between the output signal and the first input and a first P—N junction element operably coupled in a forward bias direction between the first input and a ground. The voltage reference circuit further includes a second resistance element operably coupled between the output signal and the second input, a third resistance element operably coupled to the second input, and a second P—N junction element operably coupled in series with the third resistance element, in a forward bias direction, between the third resistance element and the ground. In addition, the voltage reference circuit includes a fourth resistance element operably coupled between the second input and the ground.  
         [0019]     In another embodiment of the invention, a voltage reference circuit comprises an amplifier having a first input, a second input, and a comparison result configured as an output signal. The voltage reference circuit further includes a first resistance element operably coupled between the output signal and the first input and a first P—N junction element operably coupled in a forward bias direction between the first input and a ground. The voltage reference circuit further includes a second resistance element operably coupled between the output signal and the second input, a third resistance element operably coupled to the second input, and a second P—N junction element operably coupled in series with the third resistance element, in a forward bias direction, between the third resistance element and the ground. In addition, the voltage reference circuit includes a fourth resistance element operably coupled between the second input and the ground.  
         [0020]     Another embodiment of the invention comprises a method of generating a reference voltage. The method includes generating a reference current. The method also includes generating a first voltage signal related to a first portion of the reference current, wherein the first voltage is inversely related to a temperature change, and generating a second voltage signal related to a second portion of the reference current, wherein the second voltage is directly related to the temperature change. The method also includes comparing the first voltage signal to the second voltage signal to generate a comparison result and modifying the reference current with a current change related to the comparison result. Finally, the method also includes providing an output voltage related to the second voltage, wherein the output voltage is a voltage offset above a bandgap voltage and substantially independent of the temperature change.  
         [0021]     Another embodiment of the present invention comprises a semiconductor device including at least one voltage reference circuit according to an embodiment of the invention described herein.  
         [0022]     Another embodiment of the present invention comprises at least one semiconductor device fabricated on a semiconductor wafer, wherein the at least one semiconductor device includes at least one voltage reference circuit according to an embodiment of the invention described herein.  
         [0023]     Yet another embodiment in accordance with the present invention comprises an electronic system including at least one input device, at least one output device, at least one processor, and at least one memory device. The at least one memory device includes at least one voltage reference circuit according to an embodiment of the invention described herein. 
     
    
     BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS  
       [0024]      FIG. 1  is a circuit diagram of a conventional bandgap reference circuit;  
         [0025]      FIG. 2  is a graphical illustration of various voltages in the bandgap reference circuit of  FIG. 1 ;  
         [0026]      FIG. 3  is a circuit diagram of a conventional bandgap reference circuit for generating a voltage reference that is an integer multiple of the bandgap voltage;  
         [0027]      FIG. 4  is a circuit model of an embodiment of the present invention for generating a variable output voltage above the bandgap voltage;  
         [0028]      FIG. 5A  is a circuit diagram of an embodiment of the present invention for generating a variable output voltage above the bandgap voltage;  
         [0029]      FIG. 5B  is a circuit diagram of an embodiment of the present invention for generating a variable output voltage above the bandgap voltage and a variable output current;  
         [0030]      FIG. 6A  is a graphical illustration of various voltages according to the  FIG. 5A  embodiment;  
         [0031]      FIG. 6B  is a graphical illustration of various currents according to the  FIG. 5A  embodiment;  
         [0032]      FIG. 7A  is circuit diagram of another embodiment of the present invention for generating a variable output voltage above the bandgap voltage;  
         [0033]      FIG. 7B  is circuit diagram of another embodiment of the present invention for generating a variable output voltage above the bandgap voltage and a variable output current;  
         [0034]      FIG. 8  is a semiconductor wafer containing a plurality of semiconductor devices containing a voltage reference circuit according to the present invention; and  
         [0035]      FIG. 9  is a computing system diagram showing a plurality of semiconductor memories containing a voltage reference circuit according to the present invention.  
     
    
     DETAILED DESCRIPTION OF THE INVENTION  
       [0036]     The present invention in a number of embodiments includes methods and apparatuses for generating a reference voltage that is substantially temperature independent, substantially supply voltage independent, and at a voltage output above a bandgap voltage.  
         [0037]     Some circuits in this description may contain a well-known circuit configuration known as a diode-connected transistor. A diode-connected transistor is formed when the gate and drain of a Complementary Metal Oxide Semiconductor (CMOS) transistor are connected together, or when the base and collector of a bipolar transistor are connected together. For example, in the circuit shown in  FIG. 1 , the bipolar transistors  28  and  38  are connected in a diode configuration. When connected in this fashion the transistor operates with voltage to current properties similar to a p-n junction diode.  
         [0038]     Historically, voltage references corresponding to the bandgap voltage of silicon have been defined using the voltage from the base to emitter (V be ) of a bipolar junction transistor. However, any device creating a P—N junction may be used rather than a bipolar transistor, such as, for example a conventional diode or a CMOS device connected in a diode configuration. While the bandgap voltage may be obtained from a variety of devices in the various embodiments of the invention, suitable devices used to generate the bandgap voltage may be generally referred to as diodes, P—N junction elements, diode-connected CMOS transistors, and diode connected bipolar transistor. In addition, the voltage drop generated by any of these devices may be referred to using the historical V be  nomenclature.  
         [0039]      FIG. 4  illustrates a circuit model  90 , to show the theory of generating a reference voltage above the bandgap voltage that is substantially independent from temperature change. A current generator  92  is coupled to the series combination of a resistance element  94  and a negative temperature coefficient element  96 . The resistance element  94  provides a Proportion to Absolute Temperature (PTAT) voltage (also referred to as a positive temperature coefficient) to balance the negative temperature coefficient element  96 . The current generator  92  provides a reference current I ptco  (Positive Temperature Coefficient with an Offset Current) different from that of a conventional bandgap reference circuit such that the voltage on an output node  98  may be selected to be at a voltage higher than the bandgap voltage, as is explained more fully below.  
         [0040]      FIG. 5A  illustrates an embodiment of the present invention for generating a variable output voltage above the bandgap voltage. The voltage reference circuit  100  includes a current source  105  configured as a p-channel transistor, an amplifier  140 , a first voltage generator  150 , and a second voltage generator  160 . The first voltage generator  150  includes a first P—N junction element D 1  and a first resistance element R 1 . The second voltage generator  160  includes a second P—N junction element D 2 , a second resistance element R 2 , a third resistance element R 3 , and a fourth resistance element R 4 . The first P—N junction element D 1  and second P—N junction element D 2  are configured with junction areas of relative size such that the first P—N junction element D 1  has a junction area with a relative size of one, and the second P—N junction element D 2  has a junction area that is N times the size of the first P—N junction element D 1 .  
         [0041]     In general, embodiments of the invention are described that generate a desired voltage on an output signal  130 . However, those of ordinary skill in the art will appreciate that some applications may require a current reference rather than, or in addition to, a voltage reference. In those applications, an embodiment shown in  FIG. 5B  may be used. The embodiment of  FIG. 5B  is similar to the embodiment of  FIG. 5A  with the inclusion of an optional output current source  144 , which may be used to generate an output current signal  146  that is proportional to the voltage on the output signal  130 . In the embodiment of  FIG. 5B , a simple p-channel transistor is used for generating the output current signal  146 . Those of ordinary skill in the art, will also recognize that other current sources are possible and encompassed by the scope of the invention.  
         [0042]     Similarly, those of ordinary skill in the art will recognize that the current source  105  may be configured with a variety of circuit elements, such as, for example an n-channel transistor in a source follower configuration. Also, the resistance elements may be formed using various circuit elements and connections to generate a relatively constant resistance value. Some possible resistor implementations include, for example, discrete resistors, a length of N+ doped region as a resistor element, a length of P+ doped region as a resistor element, a length of polysilicon as a resistor element, an n-channel transistor connected such that it operates in the saturation region, and a p-channel transistor connected such that it operates in the saturation region.  
         [0043]     As stated earlier, two diodes of different sizes, but with the same emitter current, will have different current densities and, as a result, slightly different voltage drops across the P—N junction. Similarly, because different current densities result in different voltage drops, the two diodes may also be selected to have the same size (i.e., N=1) and the circuit designed to provide different currents through the two diodes. Furthermore, P—N junctions have a negative temperature coefficient wherein changes in the voltage drop across the P—N junction are inversely related to changes in temperature. In other words, as temperature rises, the voltage drop across a P—N junction falls. For example, for silicon, V be  is inversely related to temperature changes at about −2.2 mV/° C. Thus, the difference in current density creates a slightly different voltage drop across the first P—N junction element D 1  relative to the second P—N junction element D 2 .  
         [0044]     In operation, the feedback on the amplifier  140  operates to develop a steady state wherein an inverting input node  141  (also referred to as a first input) and a non-inverting input node  142  (also referred to as a second input) are maintained at substantially the same voltage potential. If the inputs are not at the same potential, the amplifier  140  acts to reduce or increase the voltage on a feedback node  148  (also referred to as a comparison result). In turn, the voltage on the feedback node  148  will increase or decrease the current through the current source  105 .  
         [0045]     In analyzing the circuit of  FIG. 5A , it can be shown, and those of ordinary skill in the art will recognize, that the voltage across a diode may be expressed as approximately,  
             VD   =       (     kT   q     )     ⁢     ln   ⁡     (     I     Is   *   A       )                 (   1   )             
 
         [0046]     where k is Boltzmann&#39;s constant, which equals about 1.3806×10−23 Joules/° K, q is electron charge, which equals about 1.602×10−19 Coulombs, T is absolute temperature in ° Kelvin, I is the forward current through the diode, Is represents a reverse saturation current of the diode, and A is the area of the P—N junction. The term kT/q is often referred to as the thermal voltage (VT). Thus, at room temperature of 300° K, VT equals about 26 millivolts.  
         [0047]     As stated earlier, the feedback on the amplifier  140  operates to move the voltage of the first voltage signal  110  and the voltage of the second voltage signal  120  to substantially the same voltage. Thus, 
 
 V   be1   =V   R3   +V   be2    (2) 
 
         [0048]     VR 3  may also be referred to as ΔV be  because it represents the difference in voltage drop between the first P—N junction element D 1  and the second P—N junction element D 2 . Substituting in the diode equation, ΔV be  may be represented as,  
                     Δ   ⁢           ⁢     V   be       =       V     be   ⁢           ⁢   1       -     V     be   ⁢           ⁢   2                     =         (     kT   q     )     ⁢     ln   ⁡     (       I   ⁢           ⁢   1       Is   *   A   ⁢           ⁢   1       )         -       (     kT   q     )     ⁢     ln   ⁡     (       I   ⁢           ⁢   2       Is   *   A   ⁢           ⁢   2       )                       =       (     kT   q     )     ⁢     ln   ⁡     (       I   ⁢           ⁢   1   *   A   ⁢           ⁢   2       I   ⁢           ⁢   2   *   A   ⁢           ⁢   1       )                       (   3   )             
 
         [0049]     If resistance elements R 1  and R 2  are selected to have the same resistance, and at steady state the voltage at the first voltage signal  110  is substantially equal to the voltage at the second voltage signal  120 , then the current II will be substantially equal to the current  12 , and equation 2 may be written as,  
               Δ   ⁢           ⁢     V   be       =         kT   q     ⁢     ln   ⁡     (   N   )         =     VT   ⁢           ⁢     ln   ⁡     (   N   )                   (   4   )             
 
         [0050]     where N equals the ratio of P—N junction area between the first P—N junction element D 1  and the second P—N junction element D 2 .  
         [0051]     The voltage on the output signal  130  is the sum of the voltage drops across the first resistance element R 1  and the first P—N junction element D 1 , which may be written as, 
 
 V   out   =V   be1   +V   R1    (5) 
 
         [0052]     The current  12  equals the sum of the sub-current  12   a  (also referred to as a first portion) and the sub-current  12   b  (also referred to as a second portion), as represented by the equation,  
               I   ⁢           ⁢   2     =         I   ⁢           ⁢   2   ⁢   a     +     I   ⁢           ⁢   2   ⁢   b       =         Δ   ⁢           ⁢     V   be         R   ⁢           ⁢   3       +       V   ⁢           ⁢   2       R   ⁢           ⁢   4                   (   6   )             
 
         [0053]     where V 2  indicates the voltage at the second voltage signal  120 . However, in a steady state, V 2  equals V be1  so equation 6 may be written as,  
               I   ⁢           ⁢   2     =         I   ⁢           ⁢   2   ⁢   a     +     I   ⁢           ⁢   2   ⁢   b       =         Δ   ⁢           ⁢     V   be         R   ⁢           ⁢   3       +       V     be   ⁢           ⁢   1         R   ⁢           ⁢   4                   (   7   )             
 
         [0054]     Therefore, the voltage drop across the second resistance element R 2  is.  
               V     R   ⁢           ⁢   2       =       R   ⁢           ⁢   2   *   I   ⁢           ⁢   2     =         (       R   ⁢           ⁢   2       R   ⁢           ⁢   3       )     ⁢   Δ   ⁢           ⁢     V   be       +       (       R   ⁢           ⁢   2       R   ⁢           ⁢   4       )     ⁢     V     be   ⁢           ⁢   1                     (   8   )             
 
         [0055]     In a steady state, V R1  equals V R2 . As a result, Vout from equation 5 may be written as,  
             Vout   =       V     be   ⁢           ⁢   1       +       (       R   ⁢           ⁢   2       R   ⁢           ⁢   3       )     ⁢   Δ   ⁢           ⁢     V   be       +       (       R   ⁢           ⁢   2       R   ⁢           ⁢   4       )     ⁢     V     be   ⁢           ⁢   1                   (   9   )             
 
         [0056]     From this equation, parameters sets may be defined that meet a voltage on the output signal  130  that is greater than the bandgap voltage of about 1.25 volts, while still maintaining substantial temperature independence wherein the change in voltage of the output signal  130  relative to a change in temperature is substantially near zero. In other words,  
           ⅆ   Vout       ⅆ   T       ≈   0       
 
         [0057]     For example, in the case of R 1 =R 2 =240 Kohms, R 3 =15 Kohms, R 4 =400 Kohms, and N=8, a Vout of about 2.2V can be obtained.  
         [0058]     In contrast, analyzing the prior art circuit of  FIG. 1 , yields an equation for the current  12 , which may be represented as,  
               I   ⁢           ⁢   2     =       Δ   ⁢           ⁢     V   be         R   36               (   10   )             
 
         [0059]     Therefore, the voltage drop across the resistance element  22  is,  
               V   22     =         R   22     *   I   ⁢           ⁢   2     =       (       R   32       R   36       )     ⁢   Δ   ⁢           ⁢     V   be                 (   11   )             
 
         [0060]     Thus, in a steady state and with V 22  equal to V 32 , the Vout of  FIG. 1  may be written as,  
             Vout   =       V     be   ⁢           ⁢   1       +       (       R   32       R   36       )     ⁢   Δ   ⁢           ⁢     V   be                 (   12   )             
 
         [0061]     In other words, Vout for the prior art circuit of  FIG. 1  may be written as Vout=V be1 +A*V be . Whereas, in embodiments of the present invention, Vout may be written as Vout=V be1 +B*ΔV be +C*V be1 .  
         [0062]     Equation 9 may be illustrated graphically by  FIG. 6A . In  FIG. 6A , line  125  illustrates the negative temperature coefficient of the first P—N junction element D 1  (i.e., the first voltage signal  110  and, in a steady state, the second voltage signal  120 ). Line  135  illustrates the voltage difference across R 2 , which is equal to the resistance of R 2  times Iptco (i.e., R 2 *Iptco). Line  135  includes a slope similar to that of  FIG. 2 , namely the (R 2 /R 3 )*ΔV be  term from equation 9. However, in  FIG. 6A , line  135  includes a y-intercept higher than that of  FIG. 2 . The y-intercept may be represented by the portion of equation 9 defined as (R 2 /R 4 )*V be1 . Line  145  represents the Vout voltage, which is a sum of line  125  and line  135 .  
         [0063]     Similarly, the current  12  may be represented graphically as in  FIG. 6B . Current I 2  is illustrated as the sum of sub-current I 2   a  and sub-current I 2   b.  It can be seen that current I 2   a  is directly related to temperature change due to the ΔV be  term in equation 7. Similarly, sub-current I 2   b  is inversely related to temperature change due to the V be1  term in equation 7. As a result, it can be seen how the current generator  92  (shown in  FIG. 4 ) can create a reference current Iptco with a positive temperature coefficient from the I 2   a  portion of current I 2  and an additional offset current from the I 2   b  portion of current I 2 .  
         [0064]     In operation of the voltage reference circuit of  FIG. 5A , the feedback on the amplifier  140  operates to develop a steady state wherein the inverting input node  141  and the non-inverting input node  142  are maintained at substantially the same voltage potential. If the inputs are not at the same potential, the amplifier  140  acts to reduce or increase the voltage on the feedback node  148 . In turn, the voltage on the feedback node  148  will increase or decrease the current through the current source  105 . Thus, for a circuit wherein the first resistance element R 1  and the second resistance element R 2  have the same value, the voltage drop across the first P—N junction element D 1  is equal to the voltage drop across the circuit combination of the second P—N junction element D 2 , the third resistance element R 3 , and the fourth resistance element R 4 . As stated earlier, due to the negative temperature coefficient for diodes, as temperature rises, the V be  of the first P—N junction element D 1  decreases at a higher rate than the V be  decrease of the second P—N junction element D 2 . Consequently, to keep the feedback loop in a steady state, the ΔV be  across the third resistance element R 3 , has a direct temperature correlation (i.e., voltage change increases as temperature increases).  
         [0065]     However, with embodiments of the present invention, the fourth resistance element R 4  provides a shunting current path to ground around the third resistance element R 3  and the second P—N junction element D 2 . This operates to increase the current I 2 , resulting in a larger voltage drop across the second resistance element R 2 . In other words, when the proper resistance ratios are selected, V 2  may be held substantially near the thermal voltage by adjusting the ratio of R 3  relative to R 2 . However, at the same time, adjusting R 4  relative to R 2  may generate a larger voltage drop across the first resistance element R 1  and the second resistance element R 2  to raise the reference voltage on the output signal  130 . Different resistance ratios may be selected to modify the reference voltage to different values while still maintaining a substantial independence from source voltage and a substantial independence from temperature changes.  
         [0066]      FIG. 7A  illustrates another embodiment of the present invention for generating a variable output voltage above the bandgap voltage. The voltage reference circuit  100  includes an amplifier  140 ′, a first resistance element R 1 ′, a second resistance element R 2 ′, a first voltage generator  150 ′, and a second voltage generator  160 ′. The first voltage generator  150 ′ comprises a first P—N junction element D 1 ′. The second voltage generator  160 ′ includes a second P—N junction element D 2 ′, a third resistance element R 3 ′, and a fourth resistance element R 4 ′. The first P—N junction element D 1 ′ and second P—N junction element D 2 ′ are configured with junction areas of relative size such that the first P—N junction element D 1 ′ has a junction area with a relative size of one, and the second P—N junction element D 2 ′ has a junction area that is N times the size of the first P—N junction element D 1 .  
         [0067]     The embodiment of  FIG. 7A  operates similar to the embodiment of  FIG. 5A  except that the output of the amplifier  140 ′ acts directly as a current source for currents I 1 ′ and I 2 ′, rather than buffering the output of the amplifier  140 ′ through a current source. In addition, the output of the amplifier  140 ′ acts as the output signal  130 ′. In operation, the explanation for the embodiment of  FIG. 5A  is equally applicable to the embodiment of  FIG. 7A .  
         [0068]     In those applications where a current reference may be desired, an embodiment shown in  FIG. 7B  may be used. The embodiment of  FIG. 7B  is similar to the embodiment of  FIG. 7A  with the inclusion of an optional output current source  144 ′, which may be used to generate an output current signal  146 ′ that is proportional to the voltage on the output signal  130 ′.  
         [0069]     Embodiments of the present invention, while mostly described in relation to semiconductor memories, are applicable to many semiconductor devices. By way of example, any semiconductor device requiring a voltage reference above the bandgap voltage, which is substantially temperature independent, such as sense amplifiers, input signal level sensors, phase locked loops, and delay locked loops, may use the present invention.  
         [0070]     As shown in  FIG. 8 , a semiconductor wafer  400 , in accordance with the present invention, includes a plurality of semiconductor devices  200  incorporating at least one embodiment of the voltage reference circuits  100  described herein. Of course, it should be understood that the semiconductor devices  200  may be fabricated on substrates other than a silicon wafer, such as, for example, a Silicon On Insulator (SOI) substrate, a Silicon On Glass (SOG) substrate, and a Silicon On Sapphire (SOS) substrate.  
         [0071]     As shown in  FIG. 9 , an electronic system  500 , in accordance with the present invention, comprises an input device  510 , an output device  520 , a processor  530 , and a memory device  540 . The memory device  540  comprises at least one semiconductor memory  200 ′ incorporating at least one embodiment of the voltage reference circuits  100  described herein in a DRAM device. It should be understood that the semiconductor memory  200 ′ might comprise a wide variety of devices other than a DRAM, including, for example, Static RAM (SRAM) devices, and Flash memory devices.  
         [0072]     While the present invention has been described herein with respect to certain preferred embodiments, those of ordinary skill in the art will recognize and appreciate that it is not so limited. Rather, many additions, deletions, and modifications to the preferred embodiments may be made without departing from the scope of the invention as hereinafter claimed. In addition, features from one embodiment may be combined with features of another embodiment while still being encompassed within the scope of the invention as contemplated by the inventors.