Abstract:
A match-insensitive low current bias circuit uses a transistor arrangement which takes advantage of the transistors&#39; collector current degeneration, current gain through emitter sizing, and voltage gain to minimize any errors caused by stage mismatches created during production. The bias circuit of the present invention is particularly suited to integrated circuit applications where a low biasing current is required.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to low current bias circuits. In particular, the present invention relates to low bias current sources in integrated circuit applications. 
     2. Description of the Related Art 
     Biasing techniques which are used in discrete circuit applications are not normally suited for use in integrated circuits. In an integrated circuit (“IC”), large resistors and capacitors are more difficult to manufacture than transistors. Consequently, IC designers have devised biasing techniques which use transistors wherever possible. In an IC, a constant current is often generated at one location and is distributed throughout the IC using current mirrors and steering circuits. 
     Biasing in IC design is often based on the well-known bandgap reference. A bandgap reference circuit takes advantage of a very stable delta base-to-emitter voltage (V BE ) between two conducting bipolar junction transistor (“BJT”) to provide a constant current, which is then used as a reference current. In one such reference current, the voltage difference (ΔV BE , typically approximately 60 mV) between two bipolar transistors&#39; V BE &#39;s are applied across a known resistance to create a reference current. The reference current is then scaled by bias circuits to bias other circuits in the IC. To ensure that the reference current is stable across the integrated circuit, the bias circuits are fabricated within a set of tolerance and specifications matching those of the bandgap reference. For example, a bias circuit designed to operate with a 2 uA current source must be coupled to a bandgap reference which can accurately provide such a current. 
     In general, transistors fabricated on the same substrate can have matched characteristics which track changes in both the fabrication process and operating parameters (e.g., temperature). Manufacturing tolerances and design tolerances determine how closely circuits can be matched. If the design is sensitive to mismatches, manufacturing tolerance must be tightened. Otherwise, low production yield and device reliability would result. Circuit matching becomes more critical as bias currents reach the sub-nanoampere level, which is required in today&#39;s power devices. 
     The following equation relates in a BJT a change in voltage V BE  to a change in collector current:                  I   new       I   old       =            Δ                   V   BE         V   T                 (   1   )                                
     where I old  and I new  are the collector currents of a BJT before and after an increase of ΔV BE  in voltage V BE ; and V T  (˜26 mV) is the thermal voltage. Equation (1) can be rewritten as:                Δ                   V   BE       =       V   T        ln          I   new       I   old                 (   2   )                                
     Thus, equation (1) provides that a 60 mV change in V BE  results in a ten-fold increase in collector current. Similarly, equation (2) provides that an 8% change in collector current results in a 2 mV change in V BE . 
     A low-current bias circuit  100  in the prior art is shown in FIG.  1 . As shown in FIG. 1, circuit  100  includes transistors Q 8  and Q 9  of equal size, and resistor R 5  (180 KΩ) coupled between an output terminal of current source  101  (which has a current I source  of 1 μA) and the collector terminal (V 5 ) of transistor Q 8 . The base terminal of transistor Q 8  is also coupled to the output terminal of current source  101 . The base terminal of transistor Q 9  is coupled to collector terminal (V 5 ) of transistor Q 8 . The collector terminal of transistor Q 9  is coupled to the circuit intended to be biased. 
     For our purpose, the base current of a BJT is negligible relative to the collector current. Thus, collector current I c8  of transistor Q 8  is equal to current I source  of current source  101 . Since resistor R 5  provides a voltage drop of 180 mV from supply voltage V CC , the V BE  of transistor Q 8  exceeds the V BE  of transistor Q 9  by 180 mV, thus output current I out  of transistor Q 9  is approximately 1 nA, as provided by equation (1) above (i.e. I out =10 −6 *e −180/26 =0.984*10 −9 ). Circuit  100  can thus be used to supply a low bias current in an IC. Also, if circuit  100  is fabricated on the same substrate as the bandgap reference circuit which provides current source  101 , circuit  100  tracks the bandgap reference over variations in fabrication process and temperature. 
     Circuit  100 , however, is sensitive to circuit mismatches. For example, if the resistance of resistor R 5  is lowered by 10% due to a variation in the fabrication process, the voltage across resistor R 5  decreases by 18 mV, which causes an increase of the same magnitude in the V BE  voltage of transistor Q 9 . Consequently, the output current I out  of transistor Q 9  doubles. Thus, a 10% change in resistor R 5  results in a 100% increase in output current I out . Clearly, such match-sensitivity does not meet today&#39;s production yield and device reliability requirements. 
     Thus, a need for a low-current bias circuit that is relatively insensitive to circuit mismatches is desired. 
     SUMMARY OF THE INVENTION 
     The present invention provides a low-current bias circuit which is relatively insensitive to circuit mismatches. In one embodiment, a circuit of the present invention combines the effects of current degeneration, current gain, and voltage gain to minimize any errors caused by circuit mismatches created during fabrication. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The present invention may be better understood, and its numerous objects, features, and advantages made apparent to those skilled in the art by referencing the accompanying drawings. 
     FIG. 1 shows a current bias circuit  100  in the prior art. 
     FIGS. 2 a  to  2   b  show current bias circuits  200  and  250 , which illustrate different aspects of a circuit of the present invention. 
     FIG. 2 c  shows a circuit  280 , which is an embodiment of the present invention. 
     FIG. 3 shows a spreadsheet for selecting component values in one embodiment. 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     To facilitate comparison between elements of the various figures, and to simplify the detailed description below, like elements in the various figures are provided like reference symbols or numerals. 
     FIG. 2 a  shows a current bias circuit  200 . Current bias circuit  200  includes transistors Q 1 , Q 2  and Q 3  of equal size, and a current source  201 . Current source  201  is coupled between supply voltage V CC  and the commonly-connected collector and base terminals of transistors Q 1  and Q 2 . The base terminal of transistor Q 3  is coupled to the collector terminal of transistor Q 2 , and the collector terminal of transistor Q 3  is coupled to supply voltage V CC . The emitter terminals of transistors Q 1 , Q 2 , and Q 3  are coupled to a ground voltage reference. 
     In circuit  200 , since transistors Q 1  and Q 2  have the same size, and their respective V BE &#39;s are the same, the current I source  (˜2 μA) of current source  201  is equally divided between the respective collector currents I 1  and I 2  of transistors Q 1  and Q 2 . (For our purpose, the base current of a BJT is negligible relative to the collector current). Thus, collector current I 2  of transistor Q 2  is approximately 1 uA. Since transistor Q 3  mirrors the current of transistor Q 2 , collector current I out  of transistor Q 3  also equals 1 uA. 
     Circuit  250  of FIG. 2 b  is substantially the same as circuit  200  of FIG. 2 a,  except that transistor Q 2  of circuit  200  is replaced in circuit  250  by transistor Q 4 , which is 10 times the size of transistor Q 1 ; also, resistor R 1  (60 KΩ) is present in circuit  250 . Resistor R 1  is coupled between the emitter terminal of transistor Q 4  and the ground reference. The size of transistor Q 4  and the resistance of resistor R 1  are selected so that collector current I 2  remains at approximately 1 uA. As can be seen from equation (1), a decrease of 60 mV in V BE  of transistor Q 4  results in a 10-fold decrease in I 2,  thus transistor Q 4  is sized to be 10 times the size of transistor Q 1  to offset the decrease in V BE  in transistor Q 4 . Thus, the resistance of resistor R 1  is selected to be 60 KΩ, to result in a voltage drop of approximately 60 mV. Since transistor Q 3  mirrors the current of transistor Q 4 , the collector current I out  of transistor Q 3  remains at 1 uA. 
     FIG. 2C shows circuit  280 , which is an embodiment of the present invention. Circuit  280  is substantially the same as circuit  250  of FIG. 2 b,  except that a 180 KΩ resistor R 2  is coupled between the output terminal of current source  201  and the collector terminal of transistor Q 4 . Since collector current I 2  of transistor Q 4  is 1 uA, the voltage across R 2  is 180 mV. Consequently, the V BE  of transistor Q 3  is 180 mV less than the V BE  of transistor Q 4 , so that a 1000 times decrease in the collector current I out  of transistor Q 3  results. In this case, current I out  becomes approximately 1 nA (1 uA/1000). Thus, circuit  280  of FIG. 2C provides a 1 nA bias current. 
     Low-current bias circuit  280  is relatively insensitive to circuit mismatches. For example, if the current I source  of current source  201  is 8% lower than 2 uA, an 8% change in collector current I 1  of transistor Q 1  results, which represents a 2 mV decrease in V BE  for transistor Q 1 , according to equation (2) above. Since the V BE  of transistor Q 1  is equal to the V BE  of transistor Q 4  plus the voltage drop V 1  across resistor R 1 , the 2 mV decrease in V BE  of transistor Q 1  is divided between the V BE  of transistor Q 4  and the voltage drop across resistor R 1 . Thus, in this example, because of R 1 &#39;s resistance and the size and the gain of transistor Q 4 , a decrease of 1 mV each is seen in the V BE  of transistor Q 4  and the voltage across resistor R 1 , and a net increase of 1 mV is seen at the collector terminal V 2  of transistor Q 4 , which is coupled to the base terminal of transistor Q 3 . Thus, the V BE  of transistor Q 3  is also increased by 1 mV, which results in a 4% increase in output current I out  of transistor Q 3 . Therefore, unlike a prior art circuit (e.g., circuit  100  of FIG.  1 ), which output current I out  varies by 100% for a 10% decrease in reference current I source , circuit  280  of FIG. 2C provides a much more stable output current. 
     The component values shown for circuit  280  of FIG. 2C are chosen for illustration purposes only. For any given application, components values and device ratios are chosen according to the invention illustrated above, and the constraints then prevailing. Component values can be affected, for example, by available die space and tolerance limits. 
     To select component values for circuit  280  of FIG. 2 c,  a designer would first set the most constricted parameter. In this case, the output and source currents are likely to be chosen first. The resistance of resistor R 2  is then selected to provide a V BE  of transistor Q 3  that would produce the desired output current. Initially, resistor R 1  is selected to provide transistor Q 4  a voltage gain of 3. For example, the resistance of resistor R 1  is selected to be 60 KΩ, if resistor R 2  is selected to be 180 KΩ. The size of transistor Q 4  can then be selected such that the resulting current gain from transistor Q 1  offsets the degeneration which results from the voltage drop across resistor R 1 , so as to result in substantially the same collector currents in transistors Q 1  and Q 4 . For example, transistor Q 4  is made 10 times larger than transistor Q 1 , if resistor R 1  is selected to be 60 KΩ and the expected collector current in transistor Q 4  is 1 uA. Similarly, transistor Q 4  can be made 100 times larger than transistor Q 1  if resistor R 1  is selected to be 120 KΩ and the expected collector current of transistor Q 4  is 1 uA. 
     After initial component values are selected, the designer can then adjust the component values to match specific requirements or design changes. For example, if output current I out  is adjusted, resistor R 2  is adjusted such that the degeneration on the V BE  of transistor Q 3  produces the desired output current. The resistance of resistor R 1  and the size of transistor Q 4  are then accordingly adjusted. Computer-aided design software is available to assist in the design process. For example, circuit simulation program SPICE and Microsoft Excel spreadsheets can be used. The use of computerized design tools is advantageous, since transcendental equations are often involved which solutions are obtained using numerical methods. Further, the interdependence of component values requires all values adjusted to be consistent with each other. FIG. 3 shows a sample Microsoft Excel ver. 5.0a spreadsheet which can be used to select component values for circuit  280  of FIG. 2 c.    
     The above detailed description is provided to illustrate the specific embodiments of the present invention and is not intended to be limiting. Numerous variations and modifications within the scope of the present invention are possible. The present invention is set forth in the following claims.