Abstract:
A generalized active reset switching network using a small choke, a pair of switches, and a capacitor is revealed. The application of the generalized active reset switching network to any of a wide variety of hard switching power converter topologies yields equivalent power converters with zero voltage switching properties, without the requirement that the magnetizing current in the main power choke be reversed during each switching cycle. In the subject invention the energy required to drive the critical zero voltage switching transition is provided by the small choke that forms part of the generalized active reset switching network. The application of the generalized active reset switching network to buck, boost, buck boost, Cuk, and SEPIC converters is shown. A variation of the generalized active reset switching network which adds a single diode to clamp ringing associated with the parasitic capacitance of off switches is also revealed.

Description:
This invention was revealed in Disclosure Document Nr. 460,697 filed Aug. 16, 1999. 
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The subject invention generally pertains to electronic power conversion circuits, and more specifically to high frequency, switched mode power electronic converter circuits. 
     2. Description of Related Art 
     There are some power conversion circuits which accomplish higher efficiencies by implementing a mechanism that accomplishes switching at zero voltage. Power loss in a switch is the product of the voltage applied across the switch and the current flowing through the switch. In a switching power converter, when the switch is in the on state, the voltage across the switch is zero, so the power loss is zero. When the switch is in the off state, the power loss is zero, because the current through the switch is zero. During the transition from on to off, and vice versa, power losses can occur, if there is no mechanism to switch at zero voltage or zero current. During the switching transitions, energy losses will occur if there is simultaneously (1) non-zero voltage applied across the switch and (2) non-zero current flowing through the switch. The energy lost in each switching transition is equal to the time integral of the product of switch voltage and switch current. The power losses associated with the switching transitions will be the product of the energy lost per transition and the switching frequency. The power losses that occur because of these transitions are referred to as switching losses by those people who are skilled in the art of switching power converter design. In zero voltage switching converters the zero voltage turn off transition is accomplished by turning off a switch in parallel with a capacitor and a diode when the capacitor&#39;s voltage is zero. The capacitor maintains the applied voltage at zero across the switch as the current through the switch falls to zero. In the zero voltage transition the current in the switch is transferred to the parallel capacitor as the switch turns off. 
     The zero voltage turn on transition is accomplished by discharging the parallel capacitor using the energy stored in a magnetic circuit element, such as an inductor or transformer, and turning on the switch after the parallel diode has begun to conduct. During the turn on transition the voltage across the switch is held at zero, clamped by the parallel diode. The various zero voltage switching (ZVS) techniques differ in the control and modulation schemes used to accomplish regulation, in the energy storage mechanisms used to accomplish the zero voltage turn on transition, and in a few cases on some unique switch timing mechanisms. 
     One of the ZVS techniques uses an inductor or transformer with relatively low inductance so that the inductor current reverses sign during each switching cycle. An example of a buck converter with this property is shown in FIG.  1  and its wave forms are illustrated in FIG.  2 . One advantage of this technique is that the switching transitions are all zero voltage transitions driven by the stored energy and current in the inductor. Another advantage is that the inductor can be made small and the inductance needs to be small in order that the current can be reversed during each switching cycle. The disadvantages are that the output current reverses each cycle so that the output capacitor must be relatively large and must store a substantial amount of energy and be able to accommodate the large ripple currents. Although the inductor can be made smaller because the inductance is reduced, the size reduction of the inductor is not as large as might be suggested by the reduction in inductance value. In a typical hard switching buck converter the output choke would be saturation limited. Its core losses would be small by comparison to its copper losses. With a small value inductor with large current swings the inductor will more likely be core loss limited, so that the cross section, the core gap, and the number of turns would need to be increased to reduce the flux swing and associated core losses. Also, in the typical hard switching buck converter in which the inductor current has a large DC component and a small AC component the AC copper winding losses are typically very small. In the FIG. 1 circuit the issue of AC winding losses must be addressed by suitable magnetic circuit element design (Litz wire or properly placed and oriented copper foil or strip) or AC winding losses will be substantial. Another disadvantage of the small inductance value technique is that there will be much higher peak currents in the choke winding and in the switches which will result in additional conduction losses in those elements. Another disadvantage of the small inductance value technique is that the energy and current available to drive the zero voltage transitions decreases as the load current increases so that in an over load condition there may be no energy available to drive a zero voltage transition and there may be substantial switching losses at the same time that the conduction losses are at their highest levels. In general, almost any power converter can be made to have zero voltage switching by this mechanism. That is, almost any power converter can be designed so that the current in its principal magnetic circuit element(s) reverses each cycle so that the stored energy in its magnetic storage element(s) is directed in a way which will enable a zero voltage transition on every switching transition. 
     OBJECTS AND ADVANTAGES 
     An object of the subject invention is to provide a power converter which is relatively simple and is capable of delivering high output power at high efficiencies and high switching frequencies. 
     Another object is to provide a converter design with minimal snubber requirements and superior EMI performance. 
     Another object is to provide a simple resonant transition converter design that can be readily used with the single frequency pulse width modulated controller integrated circuits. 
     Another object is to provide a resonant switching transition mechanism which can be designed to provide zero voltage switching over the full range of line voltage and load conditions. 
     Another object is to provide a generalized resonant switching mechanism that can be applied to a wide variety of simple non-isolated and isolated converter topologies. 
     Another object is to provide a high power conversion scheme with reduced conduction losses. 
     Another object is to provide a high frequency soft switching converter with low output filter capacitor requirements. 
     Further objects and advantages of my invention will become apparent from a consideration of the drawings and ensuing description. 
     These and other objects of the invention are provided by a novel circuit technique that uses a generalized active reset switching cell consisting of two switches, a reset capacitor, and a small resonator choke. The critical zero voltage switching transitions are accomplished using the stored magnetic energy in the small resonator choke. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The present invention is illustrated by reference to the drawings. 
     FIG. 1 illustrates a circuit schematic drawing of a prior art zero voltage switching buck converter in which the inductor current is reversed each cycle in order to provide a properly directed current for driving a zero voltage switching transition. 
     FIG. 2 illustrates the switch timing and current wave forms of the FIG. 1 circuit. 
     FIG. 3 illustrates the generalized active reset zero voltage switching cell of the subject invention. 
     FIG. 4 illustrates a generalized single main choke converter using the generalized active reset switching cell of FIG.  3 . 
     Table  1  indicates how the terminals of the FIG. 4 circuit are connected to from buck, boost, and buck boost converters. 
     FIG. 5 illustrates the FIG. 4 circuit with the terminals connected to form a buck converter. 
     FIG. 6 illustrates the FIG. 4 circuit with the terminals connected to form a boost converter. 
     FIG. 7 illustrates the FIG. 4 circuit with the terminals connected to form a buck boost converter. 
     FIG. 8 illustrates the generalized active reset switching cell augmented by a rectifier whose purpose is to clamp ringing associated with the small inductor. 
     FIG. 9 illustrates a generalized single main choke power converter using the generalized active reset switching cell of FIG.  8 . 
     FIG. 10 illustrates the circuit of FIG. 9 with its terminals connected to form a buck converter. 
     FIG. 11 illustrates a buck implementation of the subject invention. 
     FIG. 12 illustrates switch and inductor current wave forms of the FIG. 11 circuit. 
     FIG. 13 illustrates an initial condition and on state of the FIG. 11 circuit. 
     FIG. 14 illustrates a first phase of a turn off transition of the FIG. 11 circuit. 
     FIG. 15 illustrates a second phase of a turn off transition of the FIG. 11 circuit. 
     FIG. 16 illustrates a third phase of a turn off transition of the FIG. 11 circuit. 
     FIG. 17 illustrates the off state of the FIG. 11 circuit. 
     FIG. 18 is another illustration of the off state of the FIG. 11 circuit. 
     FIG. 19 illustrates a first phase of a turn on transition of the FIG. 11 circuit. 
     FIG. 20 illustrates a second phase of a turn on transition of the FIG. 11 circuit. 
     FIG. 21 illustrates a third phase of a turn on transition of the FIG. 11 circuit. 
     FIG. 22 illustrates a fourth phase of a turn on transition of the FIG. 11 circuit. 
     FIG. 23 illustrates a fifth phase of a turn on transition of the FIG. 11 circuit. 
     FIG. 24 illustrates an embodiment of the FIG. 11 circuit in which the S 1  and S 2  switches are implemented using power mosfets and the S 3  switch is implemented with a diode rectifier. 
     FIG. 25 illustrates an embodiment of the FIG. 11 circuit in which all three switches are implemented with power mosfets and augmented by a diode to clamp ringing associated with the small inductor and the parasitic capacitance of the third switch. 
     FIG. 26 illustrates the FIG. 25 circuit augmented by an LC tank circuit that provides a speed up mechanism for the switching transitions. 
     FIG. 27 illustrates the FIG. 25 circuit with its terminals rearranged to form a boost converter. 
     FIG. 28 illustrates the FIG. 25 circuit with its terminals rearranged to form a buck boost converter. 
     FIG. 29 illustrates a Cuk implementation of the subject invention. 
     FIG. 30 illustrates the switch current wave forms of the FIG. 29 circuit. 
     FIG. 31 illustrates the inductor current wave forms of the FIG. 29 circuit. 
     FIG. 32 illustrates an initial condition and on state of the FIG. 29 circuit. 
     FIG. 33 illustrates a first phase of the off transition of the FIG. 29 circuit. 
     FIG. 34 illustrates a second phase of the off transition of the FIG. 29 circuit. 
     FIG. 35 illustrates a third phase of the off transition of the FIG. 29 circuit. 
     FIG. 36 illustrates the off state of the FIG. 29 circuit. 
     FIG. 37 is another illustration of the off state of the FIG. 29 circuit. 
     FIG. 38 illustrates a first phase of the turn on transition of the FIG. 29 circuit. 
     FIG. 39 illustrates a second phase of the turn on transition of the FIG. 29 circuit. 
     FIG. 40 illustrates a third phase of the turn on transition of the FIG. 29 circuit. 
     FIG. 41 illustrates a fourth phase of the turn on transition of the FIG. 29 circuit. 
     FIG. 42 illustrates a fifth phase of the turn on transition of the FIG. 29 circuit. 
     FIG. 43 illustrates an embodiment of the FIG. 29 circuit in which the three switches are implemented using power mosfets. 
     FIG. 44 illustrates an embodiment of the FIG. 29 circuit in which the third switch is implemented with a diode and the circuit is augmented by another diode to clamp ringing associated with the small inductor and the circuit&#39;s parasitic capacitance. 
     FIG. 45 illustrates a SEPIC implementation of the FIG. 29 circuit. 
     FIG. 46 illustrates a SEPIC implementation of the FIG. 29 circuit with a clamp diode. 
     FIG. 47 illustrates a Cuk implementation with a tank circuit to speed the switching transitions. 
     FIG. 48 illustrates a Cuk implementation with the two main inductors coupled on a common core. 
     FIG. 49 illustrates a SEPIC implementation with a coupled inductor replacing the second main choke to provide isolation. 
     FIG. 50 illustrates a transformer coupled Cuk implementation of the subject invention. 
     FIG. 51 illustrates the switch current wave forms of the FIG. 50 circuit. 
     FIG. 52 illustrates the inductor current wave forms of the FIG. 50 circuit. 
     FIG. 53 illustrates the on state and the initial condition of the FIG. 50 circuit. 
     FIG. 54 illustrates the first phase of the turn off transition of the FIG. 50 circuit. 
     FIG. 55 illustrates the second phase of the turn off transition of the FIG. 50 circuit. 
     FIG. 56 illustrates the third phase of the turn off transition of the FIG. 50 circuit. 
     FIG. 57 illustrates the off state of the FIG. 50 circuit. 
     FIG. 58 is another illustration of the off state of the FIG. 50 circuit. 
     FIG. 59 illustrates the first phase of the turn on transition of the FIG. 50 circuit. 
     FIG. 60 illustrates the second phase of the turn on transition of the FIG. 50 circuit. 
     FIG. 61 illustrates the third phase of the turn on transition of the FIG. 50 circuit. 
     FIG. 62 illustrates the fourth phase of the turn on transition of the FIG. 50 circuit. 
     FIG. 63 illustrates the fifth phase of the turn on transition of the FIG. 50 circuit. 
     FIG. 64 illustrates an embodiment of the FIG. 50 circuit in which all three switches are implemented using power mosfets. 
     FIG. 65 illustrates a variation of the FIG. 64 circuit which uses a diode for the third switch and is augmented with a clamp diode. 
     FIG. 66 illustrates the FIG. 65 circuit augmented with a LC tank circuit. 
     FIG. 67 illustrates the FIG. 65 circuit wherein the two main chokes are integrated on a common core. 
     
       
         
               
             
               
               
               
               
             
           
               
                   
               
               
                 Reference Numerals 
               
               
                   
               
             
             
               
                   
               
             
          
           
               
                 100 
                 DC input voltage source 
                 101 
                 node 
               
               
                 102 
                 node 
                 103 
                 lead 
               
               
                 104 
                 lead 
                 105 
                 node 
               
               
                 106 
                 node 
                 107 
                 capacitor 
               
               
                 108 
                 switch 
                 109 
                 diode 
               
               
                 110 
                 capacitor 
                 111 
                 switch 
               
               
                 112 
                 diode 
                 113 
                 node 
               
               
                 114 
                 node 
                 115 
                 lead 
               
               
                 116 
                 lead 
                 117 
                 node 
               
               
                 118 
                 inductor 
                 119 
                 node 
               
               
                 120 
                 lead 
                 121 
                 node 
               
               
                 122 
                 capacitor 
                 123 
                 switch 
               
               
                 124 
                 diode 
                 125 
                 inductor 
               
               
                 126 
                 lead 
                 127 
                 node 
               
               
                 128 
                 node 
                 129 
                 capacitor 
               
               
                 130 
                 load 
                 131 
                 capacitor 
               
               
                 132 
                 capacitor 
                 133 
                 node 
               
               
                 200 
                 DC input voltage source 
                 201 
                 node 
               
               
                 202 
                 node 
                 203 
                 capacitor 
               
               
                 204 
                 inductor 
                 205 
                 node 
               
               
                 206 
                 diode 
                 207 
                 switch 
               
               
                 208 
                 capacitor 
                 209 
                 node 
               
               
                 210 
                 diode 
                 211 
                 switch 
               
               
                 212 
                 capacitor 
                 213 
                 node 
               
               
                 214 
                 lead 
                 215 
                 lead 
               
               
                 216 
                 node 
                 217 
                 inductor 
               
               
                 218 
                 node 
                 219 
                 capacitor 
               
               
                 220 
                 node 
                 221 
                 inductor 
               
               
                 222 
                 diode 
                 223 
                 switch 
               
               
                 224 
                 capacitor 
                 225 
                 lead 
               
               
                 226 
                 node 
                 227 
                 lead 
               
               
                 228 
                 node 
                 229 
                 capacitor 
               
               
                 230 
                 load 
                 231 
                 node 
               
               
                 300 
                 DC input voltage source 
                 301 
                 node 
               
               
                 302 
                 node 
                 303 
                 capacitor 
               
               
                 304 
                 node 
                 305 
                 capacitor 
               
               
                 306 
                 switch 
                 307 
                 diode 
               
               
                 308 
                 node 
                 309 
                 diode 
               
               
                 310 
                 switch 
                 311 
                 capacitor 
               
               
                 312 
                 inductor 
                 313 
                 inductor 
               
               
                 314 
                 node 
                 315 
                 capacitor 
               
               
                 316 
                 transformer 
                 317 
                 capacitor 
               
               
                 318 
                 node 
                 319 
                 node 
               
               
                 320 
                 diode 
                 321 
                 switch 
               
               
                 322 
                 capacitor 
                 323 
                 inductor 
               
               
                 324 
                 node 
                 325 
                 capacitor 
               
               
                 326 
                 load 
               
               
                   
               
             
          
         
       
     
    
    
     SUMMARY 
     The subject invention uses a generalized active reset switching cell consisting of two switches, a capacitor, and a small inductor in a variety of converter topologies as a substitute for the main switch to form zero voltage switching converters with similar properties to the original hard switching forms of the converters, except that first order switching losses are eliminated. During the off time of each switching cycle the current in the small inductor of the generalized cell reverses direction so that there is energy available in the small inductor to drive every switching transition. 
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     FIG. 3 illustrates a generalized active reset switching cell which can be used to provide zero voltage switching to a wide variety of hard switching converter topologies. FIG. 4 illustrates a generalized single inductor power converter based on the generalized active reset switching cell which can be made to be either a buck, boost, or buck boost converter by appropriate selection of connection of the terminals. Table  1  indicates how the terminals of the FIG. 4 circuit are connected to form the buck, boost, and buck boost topologies. FIG. 5 illustrates a buck converter using the generalized active reset switching cell. FIG. 6 illustrates a boost converter using the generalized active reset switching cell. FIG. 7 illustrates a buck boost converter using the generalized active reset switching cell. FIG. 8 illustrates an improvement to the switching cell that provides a clamp for potential ringing that would occur at the junction of the diode and the inductor when switch  3  is off (open). FIG. 9 illustrates a generalized power converter based on the modified generalized switching cell of FIG.  8 . Table  1  can be used with the FIG. 9 circuit to determine how to configure the basic switching converter types. FIG. 10 illustrates a buck converter based on the modified generalized active reset switching cell. 
     Referring to FIG. 11, there is shown a series type power processing topology. The circuit employs a source of substantially DC voltage, a switching network consisting of three switches, a reset capacitor, a small resonator inductor, a main choke, a main filter capacitor, an input capacitor, and a load. For purposes of the operational state analysis, it is assumed that the reset and output filter capacitors are sufficiently large that the voltages developed across the capacitors are approximately constant over a switching interval. It is also assumed that the main choke is sufficiently large that the current in the main choke is approximately constant over a switching cycle. Also for purposes of the operational state analysis, it is assumed that the input DC voltage source has sufficiently low source impedance that the voltage developed across the input DC voltage source is approximately constant over a switching interval. It will be assumed that the parasitic capacitors that parallel the switches are small and their effects can be ignored, except during the switching transitions. It will be assumed that diodes are ideal and have no leakage and no forward voltage drop. It will finally be assumed that the power switches are ideal; that is, lossless and able to carry current in either direction. 
     Structure 
     The structure of the circuit of the subject invention is shown in FIG. 11. A positive terminal of an input source of DC potential  100  is connected to a node  101 . A negative terminal of source  100  is connected to a node  102 . A first terminal of an input capacitor  131  is connected to the node  101 . A second terminal of capacitor  131  is connected to node  102 . A lead  103  is connected to node  101  and a node  105 . A lead  104  is connected to node  102  and to a node  106 . A first terminal of a capacitor  107  is connected to node  105 . A second terminal of capacitor  107  is connected to a node  113 . A first terminal of a switch  108  is connected to node  105 . A second terminal of a switch  108  is connected to node  113 . A cathode terminal of a diode  109  is connected to node  105 . An anode terminal of diode  109  is connected to node  113 . A first terminal of a reset capacitor  132  is connected to node  106 . A second terminal of capacitor  132  is connected to a node  133 . A first terminal of a capacitor  110  is connected to node  133 . A second terminal of capacitor  110  is connected to a node  114 . A first terminal of a switch  111  is connected to node  133 . A second terminal of switch  111  is connected to node  114 . An anode terminal of a diode  112  is connected to node  133 . A cathode terminal of diode  112  is connected to node  114 . A lead  115  is connected to node  113  and to a node  117 . A lead  116  is connected to node  114  and to node  117 . A first terminal of an inductor  118  is connected to node  117 . A second terminal of inductor  118  is connected to a node  119 . A lead  120  is connected to node  106  and to a node  121 . An anode terminal of a diode  124  is connected to node  121 . A cathode terminal of diode  124  is connected to node  119 . A first terminal of a switch  123  is connected to node  121 . A second terminal of switch  123  is connected to node  119 . A first terminal of a capacitor  122  is connected to node  121 . A second terminal of capacitor  122  is connected to node  119 . A first terminal of a choke  125  is connected to node  119 . A second terminal of choke  125  is connected to a node  127 . A lead  126  is connected to node  121  and to a node  128 . A first terminal of a capacitor  129  is connected to node  127 . A second terminal of capacitor  129  is connected to node  128 . A first terminal of a load  130  is connected to node  127 . A second terminal of load  130  is connected to node  128 . 
     Operation 
     It is assumed in this analysis that the system has reached a settled operating condition. Except for the short, but finite, switching intervals there are two states of the circuit of FIG. 11, an on state and an off state. It is also assumed, for purpose of analysis, that the switching intervals between the states are approximately zero seconds and that capacitors  107 ,  110 , and  122  are small and do not contribute significantly to the operation of the converter, except during the brief switching transitions. It is also assumed that the capacitors  131 ,  132 , and  129  are large and the voltages on these capacitors are constant over a switching cycle. 
     In operation consider an initial condition, illustrated in FIG. 13, in which the switch  108  is on and the other two switches are off. Current flows through the two inductors,  118  and  125  to the load and stored energy and current in the two inductors is increasing in magnitude, as indicated in FIGS. 12 d  and  12   e.  The current wave forms of the switches are illustrated in FIGS. 12 a,    12   b,  and  12   c.  At a time determined by the control circuit the switch  108  is turned off (opened), as illustrated in FIG.  14 . During the interval illustrated by FIG. 14 capacitor  107  is charged while the capacitors  110  and  122  are discharged, due to the currents and stored energies in the inductors  118  and  125 , as the voltages at nodes  117  and  119  fall, until the diode  112  is forward biased as illustrated in FIG.  15 . After diode  112  turns on the voltage at node  117  is clamped by diode  112 , but the voltage at node  119  continues to fall until diode  124  becomes forward biased, as illustrated in FIG.  16 . Shortly after diode  124  begins to conduct switches  111  and  123  are turned on (closed), as illustrated in FIG.  17 . The circuits of FIGS. 17 and 18 represent the off state of the converter. During the off state the voltage applied to the small inductor  118  causes its current to decrease to zero and then increase in the negative direction, as illustrated in FIG.  18  and FIG. 12 d.  During the off state all of the energy stored in the inductor  118  is transferred to the capacitor  132  and back to the inductor  118  so that the energy stored in the inductor  118  is the same at the end of the off state as it was at the beginning of the off state, but the current in the inductor  118  is reversed. At the end of the off state as determined by the control circuit the switches  111  and  123  are turned off (opened) as illustrated in FIG.  19 . When switch  123  is turned off the current in inductor  125  forces the diode  124  to conduct again. When switch  111  is turned off the current in inductor  118  forces current into capacitors  107  and  110  so that capacitor  110  is charged and capacitor  107  is discharged until the diode  109  is forward biased, as illustrated in FIG.  20 . Shortly after diode  109  begins to conduct switch  108  is turned on (closed), as illustrated in FIG.  21 . The applied voltage to the inductor  118  is now large and equal to the source  100  voltage V_IN, so that the current in the small inductor  118  changes rapidly in both magnitude and direction, as illustrated in FIG.  22  and FIG. 12 d,  until the current in the inductor  118  is equal to the current in inductor  125 , at which time the current in diode  124  becomes zero and the voltage at node  119  begins to rise charging capacitor  124 , as indicated in FIG.  23 . The voltage at node  119  will rise until the voltage reaches the level of the source  100  voltage. The converter is now in the state of the initial condition as illustrated in FIG. 13, which represents the on state of the converter. During the full cycle of operation each of the three switches were turned on and off at zero voltage. 
     Related Embodiments 
     FIG. 24 illustrates an embodiment of the FIG. 11 circuit in which the switches S 1  and S 2  are implemented with power mosfets and the switch S 3  is implemented with a diode. 
     FIG. 25 illustrates an embodiment of the FIG. 11 circuit similar to the FIG. 24 circuit except that the switch S 3  is implemented with a power mosfet and a diode D 1  is added to clamp potential ringing associated with L_RES and C 3 , where C 3  is the parasitic output capacitance of S 3 . 
     FIG. 26 is another embodiment of the FIG. 11 circuit in which an LC tank circuit is added to the generalized switching cell. The tank circuit consisting of L 1  and C 1  in series provides additional energy and current for driving the switching transitions while L_RES is also providing some energy and a delay since the time required by L_RES to reverse its current is small but not zero. The additional current provided by the tank circuit reduces the size and cost of the L_RES inductor and also reduces the insertion loss associated with L_RES. The tank circuit reduces the transition time and reduces the value of L_RES thereby enabling higher effective duty cycles and enabling effective converter operation at lower line voltages. Reducing the value of the inductor L_OUT has a similar effect as adding the tank circuit and has the additional benefit of reducing the size and cost of the inductor. The value of reducing the value of L_OUT must be weighed against the cost of reducing L_OUT in additional output filter capacitance required to obtain the desired output ripple performance. 
     FIG. 27 shows another embodiment of the subject invention in which the components are arranged to form a boost converter. The operation of the generalized switching cell is identical to the buck converter, described in detail above, but the circuit is arranged so that the main choke is connected to the input&#39;s positive terminal and the main switch is connected to the negative terminal of the input, as indicated in table  1 . 
     FIG. 28 shows another embodiment of the subject invention in which the components are arranged to form a buck boost converter. The operation of the generalized switching cell is identical to the buck converter, described in detail above, but the circuit is arranged so that the main choke is connected to the input&#39;s negative terminal, which is also the output&#39;s positive terminal, as indicated in table  1 . 
     Structure 
     The structure of the circuit of the subject invention is shown in FIG. 29. A positive terminal of a source  200  of DC potential is connected to a node  201 . A negative terminal of source  200  is connected to a node  202 . A first terminal of a capacitor  203  is connected to node  201 . A second terminal of capacitor  203  is connected to a node  205 . A first terminal of a first main inductor  204  is connected to node  201 . A second terminal of inductor  204  is connected to a node  218 . A cathode terminal of a diode  206  is connected to node  205 . An anode terminal of diode  206  is connected to a node  209 . A first terminal of a switch  207  is connected to node  205 . A second terminal of switch  207  is connected to node  209 . A first terminal of a capacitor  208  is connected to node  205 . A second terminal of capacitor  208  is connected to node  209 . An anode terminal of a diode  210  is connected to node  202 . A cathode terminal of diode  210  is connected to a node  213 . A first terminal of a switch  211  is connected to node  202 . A second terminal of switch  211  is connected to node  213 . A first terminal of a capacitor  212  is connected to node  202 . A second terminal of capacitor  212  is connected to node  213 . Node  213  is connected to a lead  214 . Lead  214  is connected to a node  216 . Node  216  is connected to a lead  215 . Lead  215  is connected to node  209 . A first terminal of a small inductor  217  is connected to node  216 . A second terminal of inductor  217  is connected to node  218 . A first terminal of a capacitor  219  is connected to node  218 . A second terminal of capacitor  219  is connected to a node  220 . A lead  225  is connected to node  202 . Lead  225  is connected to a node  226 . An anode terminal of a diode  222  is connected to node  220 . A cathode terminal of diode  222  is connected to node  226 . A first terminal of a switch  223  is connected to node  220 . A second terminal of switch  223  is connected to node  226 . A first terminal of a capacitor  224  is connected to node  220 . A second terminal of capacitor  224  is connected to node  226 . A first terminal of a second main inductor  221  is connected to node  220 . A second terminal of inductor  221  is connected to a node  228 . A lead  227  is connected to node  226 . Lead  227  is connected to a node  231 . A first terminal of an output capacitor  229  is connected to node  228 . A second terminal of capacitor  229  is connected to node  231 . A first terminal of a load  230  is connected to node  228 . A second terminal of load  230  is connected to node  231 . 
     Operation 
     It is assumed in this analysis that the system has reached a settled operating condition. Except for the short, but finite, switching intervals there are two states of the circuit of FIG. 29, an on state and an off state. It is also assumed, for purpose of analysis, that the switching intervals between the states are approximately zero seconds and that capacitors  208 ,  212 , and  224  are small and do not contribute significantly to the operation of the converter, except during the brief switching transitions. It is also assumed that the capacitors  203 ,  219 , and  229  are large and the voltages on these capacitors are constant over a switching cycle. The circuit of FIG. 29 is a Cuk form of the subject invention based on the generalized active reset switching cell. 
     In operation consider an initial condition which is also the on state of the converter, illustrated in FIG. 32, in which the switch  211  is on and the other two switches are off. Current flows from the source  200  through the inductors  204  and  217  and through the switch  211 . Current also flows from the output through the inductor  221  through the capacitor  219  through the inductor  217  and through the switch  211 . During the on state the current in the switch  211  is increasing, as illustrated in FIG. 30 a,  and the currents in all three inductors are increasing as illustrated in FIGS. 31 a,    31   b,  and  31   c.  At a time determined by the control circuit the switch  211  is turned off. The current flowing in the switch  211  is now diverted into the capacitors  208  and  212 . At the time that the switch  211  is turned off the voltages at the node  216  begins to rise and the capacitor  208  begins to discharge as the capacitor  212  begins to charge. At the same time there is some discharging of the capacitor  224  as the voltage at the nodes  218  and  220  begin to rise. This condition is shown in FIG.  33 . The voltages at the nodes  216 ,  218 , and  220  continue to rise until the diode  206  becomes forward biased clamping the voltage at node  216 . This condition is illustrated in FIG.  34 . The voltage at the nodes  218  and  220  continue to rise until the diode  222  is forward biased, as illustrated in FIG.  35 . Soon after diode  222  becomes forward biased the switches  207  and  223  are turned on, as illustrated in FIG.  36 . FIG. 36 represents the off state of the converter. During the off state the current in the inductor  217  ramps down to zero then ramps up in the opposite direction to the same magnitude that it had at the beginning of the off state. This is illustrated in FIG.  37  and in FIG. 31 c.  During the off state all of the energy stored in the inductor  217  is transferred to the capacitor  203  and then the energy is transferred back to the inductor  217  so that the energy stored in the inductor  217  is the same at the end of the off state as it was at the beginning of the off state, but the current in the inductor  217  is reversed. At a time determined by the control circuit the switches  207  and  223  are turned off. The current in the inductor  217  is channeled into capacitors  208  and  212  charging capacitor  208  and discharging capacitor  212 . During this time the current in the switch  223  is diverted into the diode  222 , as illustrated in FIG.  38 . When the voltage at node  216  falls to the level of the negative terminal of source  200  the diode  210  begins to conduct, as illustrated in FIG.  39 . Soon after diode  210  begins to conduct switch  211  is turned on at zero voltage, as illustrated in FIG.  40 . At this point there is a large voltage applied across inductor  217  so that the current in the inductor  217  is changing rapidly, as indicated in FIGS. 31 c  and  30   a.  The current in the inductor  217  will change sign, as illustrated in FIG. 41, and ramp up to the level of the sum of the currents in inductors  204  and  218 . During this time interval the current in diode  222  is ramping down towards zero, as illustrated in FIG. 30 c.  When the current in the diode  222  reaches zero the voltages at the nodes  218  and  220  begins to drop as the capacitor  224  begins to charge, as illustrated in FIG.  42 . When the voltage at node  218  reaches a level near the negative terminal of the source  200  the charging of capacitor  224  is complete and the circuit enters a first on state, which is the initial condition, as illustrated in FIG.  32 . During the full cycle of operation each of the three switches were turned on and off at zero voltage. 
     Related Embodiments 
     FIG. 43 illustrates an embodiment of the FIG. 29 circuit in which all three of the switches are implemented with power mosfets. 
     FIG. 44 illustrates an embodiment of the FIG. 29 circuit similar to the FIG. 43 circuit except that the S 3  switch is implemented with a diode and a diode D 2  is added to clamp potential ringing associated with L_RES and C 3 , where C 3  is the parasitic output capacitance of D 1 . 
     FIG. 45 illustrates another embodiment of the FIG. 29 circuit in which the positions of the output choke and output switch are rearranged to form a SEPIC form of the converter, rather than the Cuk form. The differences between the Cuk form and SEPIC form are well known to those skilled in the art of power conversion. One difference is that the Cuk form yields an output that is inverted with respect to the input and the output of the SEPIC form is non-inverted. Another difference is that the SEPIC relies on the output capacitor to hold up the load when the S 3  switch is off. 
     FIG. 46 illustrates another embodiment in the SEPIC form of the invention with a clamp diode to prevent ringing of the output switch parasitic capacitance. 
     FIG. 47 illustrates another embodiment in the Cuk form of the invention with an LC tank circuit used to speed up the switching transitions and to reduce the value of the small inductor L_RES, thereby reducing the insertion loss of L_RES and enabling operation at lower line voltages. 
     FIG. 48 illustrates another embodiment of the invention in the Cuk form in which the two main chokes are coupled and integrated onto a single core. 
     FIG. 49 illustrates another embodiment of the invention in the SEPIC form in which the output inductor is replaced by a coupled inductor which provides for an output with galvanic isolation. 
     Structure 
     The structure of the circuit of the subject invention is shown in FIG. 50. A positive terminal of a DC input power source  300  is connected to a node  301 . A negative terminal of source  300  is connected to a node  302 . A first terminal of a capacitor  303  is connected to node  301 . A second terminal of capacitor  303  is connected to a node  304 . A cathode terminal of a diode  307  is connected to node  304 . An anode terminal of diode  307  is connected to a node  308 . A first terminal of a switch  306  is connected to node  304 . A second terminal of switch  306  is connected to node  308 . A first terminal of a capacitor  305  is connected to node  304 . A second terminal of capacitor  305  is connected to node  308 . A cathode terminal of a diode  309  is connected to node  308 . An anode terminal of diode  309  is connected to node  302 . A first terminal of a switch  310  is connected to node  308 . A second terminal of switch  310  is connected to node  302 . A first terminal of a capacitor  311  is connected to node  308 . A second terminal of capacitor  311  is connected to node  302 . A first terminal of an inductor  312  is connected to node  308 . A second terminal of inductor  312  is connected to a node  314 . A first terminal of an inductor  313  is connected to node  301 . A second terminal of inductor  313  is connected to node  314 . A first terminal of a capacitor  315  is connected to node  314 . A second terminal of capacitor  315  is connected to an undotted terminal of a primary winding of a transformer  316 . A dotted terminal of the primary winding of transformer  316  is connected to node  302 . A dotted terminal of a secondary winding of transformer  316  is connected to a first terminal of a capacitor  317 . An undotted terminal of the secondary winding of transformer  316  is connected to a node  319 . A second terminal of capacitor  317  is connected to a node  318 . A cathode terminal of a diode  320  is connected to node  318 . An anode terminal of diode  320  is connected to node  319 . A first terminal of a switch  321  is connected to node  318 . A second terminal of switch  321  is connected to node  319 . A first terminal of a capacitor  322  is connected to node  318 . A second terminal of capacitor  322  is connected to node  319 . A first terminal of an inductor  323  is connected to node  318 . A second terminal of inductor  323  is connected to a node  324 . A first terminal of a capacitor  325  is connected to node  324 . A second terminal of capacitor  325  is connected to node  319 . A first terminal of a load  326  is connected to node  324 . A second terminal of load  326  is connected to node  319 . 
     Operation 
     It is assumed in this analysis that the system has reached a settled operating condition. Except for the short, but finite, switching intervals there are two states of the circuit of FIG. 50, an on state and an off state. It is also assumed, for purpose of analysis, that the switching intervals between the states are approximately zero seconds and that capacitors  305 ,  311 , and  322  are small and do not contribute significantly to the operation of the converter, except during the brief switching transitions. It is also assumed that the capacitors  303 ,  315 ,  317 , and  325  are large and the voltages on these capacitors are constant over a switching cycle. The circuit of FIG. 50 is an implementation of the generalized active reset switching cell in the transformer coupled Cuk form. 
     In an initial condition illustrated in FIG. 53 the switch  310  is on and the switches  306  and  321  are off. Current is flowing from the source  300  through the inductor  313  through the inductor  312  through the switch  310  and back to the source  300 . Current also flows in a loop consisting of the primary winding of transformer  316 , the capacitor  315 , the inductor  312 , and the switch  310 . The current in the primary winding of the transformer  316  flows out of the undotted terminal. A current is induced in the secondary winding of the transformer  316  which flows out of the dotted terminal, through the capacitor  317 , through the inductor  323  to the load  326  and the output filter capacitor  325 . The initial condition also represents a first on state of the converter during which time the currents in all three inductors is increasing as illustrated in FIGS. 52 a,    52   b  and  52   c.  At a time determined by the control circuit the switch  310  is turned off, as illustrated in FIG.  54  and FIG. 51 a.  The current flowing in switch  310  is diverted to capacitors  311  and  305 . During this time the voltage at node  308  rises as capacitor  311  charges and capacitor  305  discharges. During this time the voltage at node  314  begins to rise as the voltage at node  318  begins to fall and capacitor  322  begins to discharge. The voltage at node  308  rises up until the diode  307  becomes forward biased, as illustrated in FIG.  55 . The voltage at node  314  rises up and the voltage at node  318  falls until the diode  320  becomes forward biased, as illustrated in FIG.  56 . Shortly after diode  320  becomes forward biased switches  306  and  321  are turned on at zero voltage, as illustrated in FIG.  57 . FIG. 57 represents the off state of the converter. During the off state the currents in inductors  313  and  323  are ramping down, as illustrated in FIGS. 52 a  and  52   b.  The current in inductor  312  is ramping down too, but at a much higher rate and the current in inductor  312  drops to zero, reverses, and climbs up to its magnitude at the beginning of the off state, as illustrated in FIG.  58  and FIG. 52 c.  During the off state all of the energy stored in the inductor  312  is transferred to the capacitor  303  and back to the inductor  312  so that the energy stored in the inductor  312  is the same at the end of the off state as it was at the beginning of the off state, but the current in the inductor  312  is reversed, as illustrated in FIGS. 58 and 52 c.  When the current in inductor  312  has reached its magnitude at the beginning of the off state the switches  306  and  321  are turned off, as illustrated in FIG.  59 . The current from switch  306  is diverted into capacitors  305  and  311 . The current from switch  321  is diverted into diode  320 . During this time the voltage at node  308  falls as capacitor  311  is discharged and capacitor  305  is charged. When the voltage at node  308  falls to the level of the negative terminal of source  300  diode  309  becomes forward biased, as illustrated in FIG.  60 . Soon after diode  309  turns on switch  310  is turned on at zero voltage, as illustrated in FIG.  61 . The applied voltage on inductor  312  is now large so that its current is changing rapidly, as illustrated in FIGS. 52 c,  and the current in diode  320  is also ramping down rapidly. The current in inductor  312  reverses again as indicated in FIG.  62 . When the current in diode  320  reaches zero it becomes reverse biased and the voltage at node  318  rises up charging capacitor  322 , as illustrated in FIG. 63, as the voltage at node  314  falls toward the voltage of the negative terminal of source  300 , at which time the circuit enters the on state as illustrated in FIG. 52, and a full cycle of operation has been completed. 
     Related Embodiments 
     FIG. 64 illustrates an embodiment of the FIG. 50 circuit in which all three of the switches are implemented with power mosfets. 
     FIG. 65 illustrates an embodiment of the FIG. 50 circuit similar to the FIG. 64 circuit except that the S 3  switch is implemented with a diode, D 1 , and a diode, D 2 , is added to clamp potential ringing associated with L_RES and the parasitic capacitance of D 1 . 
     FIG. 66 illustrates another embodiment in which an LC tank circuit is added to speed the switching transition and reduce the value of L_RES and the associated insertion loss of L_RES, thereby enabling circuit operation at lower line voltages. 
     FIG. 67 illustrates an embodiment in which the input and output chokes are integrated into a single coupled inductor on a common core. 
     Additional Embodiments 
     Additional embodiments are realized by applying the generalized active reset switching cell to other converter topologies. The buck, boost, buck-boost, Cuk, and SEPIC converters are shown here as examples, but it is clear to one skilled in the art of power conversion that by extending the techniques illustrated and demonstrated here to other hard switching topologies that these other hard switching topologies can be converted from hard switching converters to soft switching converters with the elimination of first order switching losses. 
     CONCLUSION, RAMIFICATIONS, AND SCOPE OF INVENTION 
     Thus the reader will see that the power converters of the invention provide a mechanism which significantly reduces switching losses, has low component parts counts, and does not require high core losses, high output filter capacitance, or high conduction losses to accomplish zero voltage switching, relying on the energy stored in a small magnetic circuit element. 
     While my above description contains many specificities, these should not be construed as limitations on the scope of the invention, but rather as exemplifications of preferred embodiments thereof. Many other variations are possible. For example, interleaved, parallel power converters with two or more parallel converter sections; power converters arranged in a bridged configuration for amplifier and inverter applications; power converters similar to those shown in the drawings but which integrate individual magnetic circuit elements onto a single magnetic core; power converters similar to those shown but which have instead high AC ripple voltages on input filter capacitors; power converters, similar to those shown in the drawings, but where the DC input source is instead a varying rectified AC signal. Accordingly, the scope of the invention should be determined not by the embodiments illustrated, but by the appended claims and their legal equivalents.