Abstract:
A communications receiver and a method for receiving and processing information transmitted on either a wide band carrier or a narrow band carrier having In-phase-Quadrature-phase (IQ) modulation, comprising, detecting a portion of the spectrum wide enough to encompass the wide band carrier (BW), converting the wide band carrier to baseband in I and Q components, each component having a bandwidth of BW/2, converting the I and Q components into further I and Q components to form components II, IQ, QI, and QQ of bandwidth equal to BW/4, where each of the sub-bands may contain a portion of the originally transmitted information. Operating in wideband mode, each of the components/is separately processed to extract portions of the originally transmitted information, and operating in a narrowband mode, each of the components containing information is separately processed within the narrow band transmitted carrier to extract portions of the originally transmitted information. The components are then recombined to reconstruct the originally transmitted information.

Description:
FIELD OF THE INVENTION 
     This invention relates to communications receivers, and more particularly to receivers capable of receiving wideband signals in a given mode such as Wide Band Code Division Multiple Access (WBCDMA) and also medium or narrow band signals for another mode such as Time Division Multiple Access (TDMA), of which the European cellular telephone system GSM (Global System Mobile) is one example. 
     BACKGROUND OF THE INVENTION 
     Present communication systems, for example cellular telephone communication systems, operate on narrow or medium bandwidth technologies. The GSM cellular telephone system, a Time Division Multiple Access (TDMA) system, is one such medium bandwidth system. As requirements for higher speed transmission for, for example, data or video transmission increase, wider band technologies are becoming more desirable. One such wideband technology for cellular communications is Wide Band Code Division Multiple Access (WBCDMA). As new technologies develop it is common to provide communications equipment capable of operating on more than one communications system to facilitate the transition from one system to the other or to allow users access to the combined capacity and features of both systems. Thus there is a need for communications receivers capable of receiving signals from both narrow or medium bandwidth systems and also from wide bandwidth communications systems. 
     Present efforts to provide a single wideband and medium band receiver have resulted in receiver designs which merely duplicate the receiver circuitry for each mode by providing different receiver data paths for each mode. As an example, in the Wideband CDMA system currently being developed in Europe, Universal Mobile Telephone System (UMTS), the channel bandwidth is 3.84 MHz while the current European cellular system, GSM, has a bandwidth of 200 KHz. Both of these systems operate in an IQ modulation mode wherein information to be transmitted, after being appropriately encoded, is provided in In-phase and Quadrature-phase modulation components superimposed on a carrier signal producing a complex signal which is subsequently demodulated and decoded by the receiver to reveal the originally encoded information. 
     Currently these two modes are combined into one communications receiver unit by providing two different receiver circuits with different downconversion mixers, blocking filters, amplifiers, antialiasing filters, and analog to digital converters. Each of the receiver circuits is thus a separate receiver capable of receiving and decoding the I and Q components of the transmitted signal. Accordingly, it would be advantageous to provide a single communications receiver capable of operating in several modes and in which many elements of circuitry are usable in each mode, thus reducing the complexity of the receiver by reducing the number of circuits necessary in the receiver. 
     Since the bandwidth of the WBCDMA UMTS system is 3.84 MHz, each I and Q component of the WBCDMA signal will be in the range of 3.84 Mhz/2. These wide bandwidths, given the required dynamic range of the transmitted signal, require the use of sophisticated data converters such as Flash or Pipelined data converters which, while quite fast thus allowing high sampling rates to accommodate code tracking to recover the transmitted code in a spread spectrum system, are not particularly suitable for the GSM mode since these data converters have high power consumption and have limited dynamic range (10 bits) while the GSM mode requires higher dynamic range (14 bits) but allows lower sampling rates. Lower power consumption, of course, is always desirable in portable or mobile communications equipment such as cellular telephones. The GSM mode allows the use of Sigma-Delta digital to analog converters or other converters which are easily programmable for different bandwidths, have wider dynamic range, and consume less power. 
     SUMMARY OF THE INVENTION 
     The above and other advantages can be obtained by superimposing the sub-bands of a wideband channel into narrower bandwidth components called II, IQ, QI, QQ using a dual low Intermediate Frequency (IF) digital approach such that each component is capable of using common data converters that can be reused when a medium or narrow bandwidth mode is selected. 
     This may be achieved by providing a dual mode communications receiver for detecting and demodulating radio signals carrying information which has been encoded and modulated onto a carrier of either wide or narrow bandwidth for transmission, comprising means for subdividing the detected band into sub-bands and superimposing the sub-bands into a plurality of components with a bandwidth similar to the sub-band bandwidth, means for processing that portion of the information contained in each component separately, and means for combining the processed information from the components to reconstruct the original information transmitted. 
     This can be accomplished as well by providing a method for operating a dual mode communications receiver for detecting and demodulating radio signals carrying information which has been encoded and modulated onto a carrier of either wide or narrow bandwidth for transmission, comprising subdividing the detected signals into detected sub-bands, superimposing the detected sub-bands into a plurality of components with a bandwidth similar to the sub-band bandwidth, processing that portion of the information contained in each component separately, and combining the processed information from the components to reconstruct the original information transmitted. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is an overall block diagram of a communications receiver in accordance with the invention. 
         FIG. 2  is a spectrum representation of the input signal Vin. 
         FIG. 3   a  and  FIG. 3   b  are spectrum representations of the input signal Vin with the real and imaginary parts separated. 
         FIG. 4   a ,  FIG. 4   b ,  FIG. 4   c , and  FIG. 4   d  are spectrum representations of the components II, IQ, QI, and QQ which are the signal Vin after mixing each of the sub-bands with sin(wt) or cos(wt) and filtering each of the components. 
         FIG. 5   a  and  FIG. 5   b  represent graphically the Vout++ vector which corresponds to the reconstruction of sub-band 0  and sub-band 1 , and the Vout−+ vector which corresponds to the reconstruction of sub-band 2  and sub-band 3 . 
         FIG. 6  is a schematic diagram of a Combiner circuit according to the instant invention. 
         FIG. 7  is an overall block diagram of a communications receiver in accordance with the invention showing the configuration of the receiver for reception of narrow or medium bandwidth signals. 
     
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENT 
       FIG. 1  is an overall block diagram of a communications receiver in accordance with the invention., An input signal Vin, which is a complex signal representing the baseband spectrum which will be translated to a Radio Frequency or Intermediate Frequency channel as earlier described, having information applied by a modulator in a transmitter to produce the input signal having In-phase and Quadrature-phase components, is applied to an amplifier  100  the output of which is applied to two quadrature mixers or downconverters  102 ,  104 . Vin has a bandwidth of BW. Vin is subdivided into four sub-bands. 
     The IQ downconverters  102 , 104  downconverts either a Radio Frequency Signal (RF) or an Intermediate Frequency (IF) channel to DC spectrum using, for example, either a Gilbert Type of Mixer or a Chopper Type of Mixer. 
     A Local Oscillator  105  drives two digital to analog converters  106  and  107  to produce quadrature analog outputs (sin(wt) and cos(wt), where w=2*Pi*BW/4) which in turn drive quadrature network  108  which divides by Pi/2 in order to deliver two RF or IF oscillator signals which are in quadrature to feed the two Local Oscillator ports of the quadrature mixers 102,104. 
     The outputs of mixers  102  and  104  then are quadrature representations of the input signal Vin at baseband or a low Intermediate Frequency (hereafter references to baseband may alternatively imply low IF, depending on receiver design). 
     The outputs of the mixers  102  and  104  are filtered by a first low pass blocking filter (capacitors) and then are applied respectively to further mixers (multipliers)  110 ,  112 ,  114 , and  116  where the I and Q sub-band components of bandwidth BW/2 are further divided into four components, II, IQ, QI, and QQ, of bandwidth BW/4 which are amplified and filtered by low pass filters  118 ,  120 ,  122 , and  124  for further blocking filtering, each having a bandwidth equal to or greater than BW/4. 
     The mixers are low frequency mixers/multipliers except the I and Q IF or RF to baseband mixers  102  and  104 . The mixers/multipliers  110 , 112 , 114  and  116  could be implemented as multiplier DAC&#39;s with gain, where the digital input port has the digital representation of cos(wt) and sin(wt) applied and the analog input port is the filtered output of the mixers  102  and  104 . The clock frequency of the multiplier DAC depends on the amount of blocking provided in front to avoid spurious response at the harmonics of this clock. Also, this multiplier DAC could be used to set an Automatic Gain Control (AGC) signal in the receiver. 
     The Low pass filters are generally based on active RC type of filter with programmable cut-off frequencies such that wide-band or medium band or narrow band can be filtered. Those filters have a bandwidth equal to BW/4 which simplifies their design versus filters of bandwidth equal to BW/2 with similar blocking rejection requirements. This allows the filters to have a reduced number of poles required per filter that then introduces less group and amplitude ripple. Also, the programming cut-off frequency range is reduced. 
     The outputs of low pass filters  118 ,  120 ,  122 , and  124  are subsequently applied to Analog to Digital converters  126 ,  128 ,  130 , and  132 , which may be Sigma-Delta A/D converters, and decimation filters  134 ,  136 ,  138 , and  140  to produce component signals IIf, IQf, QIf, and QQf which will be described in detail later. The signals are gain and phase corrected and recombined at  142  to produce output signals Vout++ and Vout−+ which are subsequently demodulated and decoded to produce a digital bit stream representing the information previously encoded onto the complex input signal by the modulator of the transmitter. 
     The Analog to Digital converters  126 ,  128 ,  130 , and  132  are preferrably Sigma-Delta type A/D modulators with a programmable oversampling ratio for various wideband or medium or narrow band signal components. Since the components II, IQ, QI, QQ have their bandwidths reduced to BW/4 each, this reduces the required oversampling clock frequency of oversampling for the Sigma-Delta for the same Dynamic Range. For example, in WBCDMA mode with BW=3.84 Mhz, the oversampling clock would be selected in the range of 26 Mhz such that the oversampling ratio is in the range of 26 Mhz/(3.84 Mhz/4)=27.08 for 10 bits of resolution, while in GSM mode, the oversampling clock would be 13 Mhz for an oversampling ratio 13 Mhz/0.2 khz=65 for 14 bits of resolution. 
     The Sigma-Delta modulators  126 , 128 , 130  and  132  generate digital streams oversampled at specified frequencies of oversampling. These digital outputs will contain shaped noise spectrum depending on the order and the type of Sigma-Delta modulator as is typical for this kind of modulator. The digital outputs are then digitally filtered to remove the shaped noise of the Sigma-Delta modulators and then decimated to a lower clock frequency for further processing by decimation digital filters  134 , 136 , 138  and  140  operating at lower clock rates to perform selectivity filtering. The digital outputs IIf, IQf, Qif, and QQf are then processed by digital multipliers and adders in Combiner  142  in order to provide gain and phase imbalance correction and to generate four digital outputs components represented by vectors Vout++ and Vout−+. 
     The bandwidths of analog filters  118 ,  120 ,  122 , and  124 , the Sigma-Delta converters  126 ,  128 ,  130 , and  132 , and the digital decimation filters  134 ,  136 ,  138 , and  140  can also be made programmable to handle several bandwidth settings. 
     In this preferred embodiment as shown in  FIG. 1  a receiver is described in which the input signal is divided into four components, but it is understood that a different number of components may be used depending upon the bandwidth of the input signal and the desired bandwidth of the components. 
     The operation of the receiver of  FIG. 1  can best be described with reference to the spectrum representations of the signals at various points in the receiver as set forth in  FIGS. 2-5 . An input channel applied to amplifier  100  is subdivided into four sub-bands hereafter referred to as sub 0 , sub 1 , sub 2 , and sub 3  and referred to by reference numerals  160 ,  162 ,  164 , and  166 , respectively, in  FIG. 2  which is a spectrum representation of Vin after having been divided into four sub-bands. Each of the sub-bands has a bandwidth of BW/4. 
     The RF channel on which those four sub-bands is based is downconverted from the RF or IF frequency to baseband in two components, I and Q, as the outputs of mixers  102  and  104 , which are in quadrature phase relationship and where each component has a spectrum width of BW/2 (from 0 to BW/2). 
     If the vector Vin=I+j.Q is the representation of the input channel at baseband, then Vin can be expressed as a sum of four vectors that correspond to the four sub-bands, e.g., Vin=Vin 0 +Vin 1 +Vin 2 +Vin 3 .  FIG. 2  also shows this representation where Vin 0  corresponds to sub 0 , Vin 1  corresponds to sub 1 , Vin 2  corresponds to sub 2 , and Vin 3  corresponds to sub 3 . 
       FIG. 3   a  and  FIG. 3   b  show the spectrum of Vin represented in its constituent I and Q components (above the lines f). Thus are shown the real part of Vin, e.g., I=I 0 +I 1 +I 2 +I 3 , shown, respectively, as  160 I,  162 I,  164 I, and  166 I in  FIG. 3   b , and the imaginary part j.Q=j.Q 0 +j.Q 1 +j.Q 2 +j.Q 3 , shown, respectively, as  160 Q,  162 Q,  164 Q, and  166 Q in  FIG. 3   b.    
     As previously noted, then sub 3 ,  166 , is mixed with sub 0 ,  160 , sub 2 ,  164 , is mixed with sub 1 ,  162 , again looking separately at the I and j.Q spectra below the f lines. 
     Each component I and Q is then mixed with a low IF quadrature signals cos(wt) and sin(wt) in mixers  110 ,  112 ,  114 , and  116  (where w=2.pi.BW/4 or a nearest value), i.e. the Low IF clock is half the I or Q bandwidth. 
     The operation is as follows:
 
 IIf=II**HII  with  II=I  cos(wt)
 
 IQf=IQ**HIQ  with  IQ=I  sin(wt)
 
 QIf=QI**HQI  with  QI=Q  cos(wt)
 
 QQf=QQ**HQQ  with  QQ=Q  sin(wt)
 
Where HII, HIQ, HQI and HQQ are the transfer function of the analog and digital filters  118 ,  120 ,  122 , and  124 , and  134 ,  136 ,  138 , and  140 , respectively, on the II, IQ, QI and QQ paths respectively.
 
     The ** denotes as a time domain convolution.
 
 II=I .( e   +j(wt)   +e   −j(wt) )/2
 
 IQ=I .( e   +j(wt)   −e   −j(wt) )/2 j 
 
 QI=Q .( e   +j(wt)   +e   −j(wt) )/2
 
 QQ=Q .( e   +j(wt)   −e   −j(wt) )/2 j 
 
     Multiplying with e +j(wt)  is equivalent to shift the spectrum by +w , i.e by +BW/4. 
     Multiplying with e −j(wt)  is equivalent to shift the spectrum by −w , i.e by BW/4. 
     The result is the spectrum shown in  FIG. 4   a ,  FIG. 4   b ,  FIG. 4   c , and  FIG. 4   d  which also graphically denotes the outputs of each of the filters  118 ,  120 ,  122 , and  124 , respectively, as II, IQ, QI, and QQ, each shown in the FIGS. As a filter of bandwidth BW/2 and the outputs of filters  134 ,  136 ,  138 , and  140  as IIf, IQf, QIf, and QQf, respectively, also shown as filters of bandwidth W/2. The spectrum shifts as described above are also shown graphically in FIGS.  4 . Note that each of the outputs of the filters in the II, IQ, QI, and QQ paths contains each of the four sub-bands, that is why the components II, IQ, QI, and QQ have the four sub-bands superimposed, as shown in  FIG. 4   a - FIG. 4   d , and thus all the information originally contained within the input signal Vin is available to reconstruct the original signal, but the processing of the signals is at narrower bandwidths than the original signal Vin and thus can be performed with Sigma-Delta A/D converters rather than the flash or pipelined data converters usually used for this operation to support wide bandwidths. 
     To produce the output in the form of the reconstruction of the original (but now processed) information transmitted, the information of each of the four sub-bands must be recombined in Combiner  142  to form two vector components Vout++ and Vout−+:
 
 V out++=( IIf−QQf )+ j .( IQf+QIf )
 
 V out−+=( IIf+QQf )+ j .(− IQf+QIf )
 
     Then the Vout++ vector corresponds to the reconstruction of sub-band 0  and sub-band 1 ,  160  and  162 , and the Vout−+ vector corresponds to the reconstruction of sub-band 2  and sub-band 3 ,  164  and  166 , as shown graphically in  FIG. 5   a  and  FIG. 5   b , respectively. The recombination by Combiner  142  can be performed in several ways, for example: 
     Use an adder/subtractor to perform IIf−QQf=I++ for Vout++. 
     Use an adder/subtractor to perform IQf+QIf=Q++ for Vout++ 
     Use an adder/subtractor to perform IIf+QQf=I−+ for Vout−+ 
     Use an adder/subtractor to perform −IQf+QIf=Q−+for Vout−+ 
     The operation Vout++e −j(wt)  is formed using a complex multiplier to perform the following digital operation: (I+++j.Q++).(cos(wt)−j.sin(wt))=I++.cos(wt)+Q++.sin(wt)+j.(−I++.sin(wt)+Q++.cos(wt)) using adders and multipliers. 
     The operation Vout−+e j(wt)  is formed using a complex multiplier to perform the following digital operation: (I−++j.Q−+).(cos(wt)+j.sin(wt))=I−+.cos(wt)−Q−+.sin(wt)+j.(I−+.sin(wt)+Q−+.cos(wt)) using adders and multipliers. 
     Note that all four of the components IIf, IQf, QIf, and QQf are required to reconstruct each sub-band. It is required that all four components be available to recover the sub-band separately. Each component (II, IQ, QI, and QQ) contains within the BW/4 information related to all the four sub-bands as shown in their respective spectrum. That is what allows the receiver to process all four sub-bands in parallel and allows the use of more favorable A/D conversion techniques. For this reason the receiver can be referred to as a Double Cartesian receiver since the four components are required to reconstruct each sub-band. 
     Note also that those four real components have a spectrum width of BW/4 and that the same low pass filters with the same low pass A/D converters are required to digitize those four components. 
     The receiver employing the instant invention could be seen as a receiver having an RF or IF input channel of bandwidth BW with four baseband output components (Double Cartesian) called IIf, QQf, QIf, and IQf that have a bandwidth requirement of BW/4. To reconstruct back the original spectrum, the two reconstructed vector Vout++ and Vout−+ needed to be shifted by −w and by +w respectively, and so are multiplied, respectively by e −j(wt−phi0)  and e +j(wt−phi0) :
 
 V out  r=V out++. e   −j(wt−phi0)   +V out−+. e   +j(wt−phi0) 
 
Where phi0 is chosen to remove phase discontinuities. Since Sin(wt) and cos(wt) need to be generated, the phase generator that generated sin(wt) and cos(wt) simply starts with an offset of −phi0.
 
     By replacing Vout++ and Vout−+ by their expressions we get:
 
 V out  r= 2( IIf .cos( wt−phi 0)+ IQf .sin( wt−phi 0))+2 j .( Qif .cos( wt−phi 0)+ QQf .sin( wt−phi 0))
 
i.e., the reconstruced In-phase and Quadrature components are:
 
 Ir =( IIf .cos( wt−phi 0)+ IQf .sin( wt−phi 0))
 
 Qr =( QIf .cos( wt−phi 0)+ QQf .sin( wt−phi 0))
 
     A reconstruction implementation of Combiner  142  is shown in  FIG. 6  to generate the real and imaginary output of the reconstructed vector Vout r in accordance with the equations set forth above and also provide gain and phase correction for mismatches introduced by the four paths. The recombination equations Ir=(IIf.cos(wt−phi0)+IQf.sin(wt−phi0)) and Qr=(QIf.cos(wt−phi0)+QQf.sin(wt−phi0)) implemented in  FIG. 6  also use gain and phase corrections between the components when they do not match among themselves. 
     IIf is mutliplied by the gain correction term Kii, IQf is multiplied by the gain correction term Kiq, QQf is multiplied by the gain correction term Kqq and QIf is multiplied by the gain correction Kqi by multipliers  170 ,  172 ,  174 , and  176 , respectively. The gain corrections terms are chosen such the corrected terms are equal in amplitude. 
     It is still necessary to correct the phase between the IIf, IQf, Qif, and QQf. Rather than multiplying, for example, IIf by cos(wt−phi0), it is multiplied by cos(wt−phi0+PhiIQ) in multiplier  178  where PhiIQ is a phase value used to phase shift IIf versus IQf to compensate the phase mismatch between IIf and IQf. Similarly, the term PhiQI is the phase correction to compensate the phase mismatch between QIf and QQf applied in multiplier  180 . 
     Once IIf and IQf are gain and phase matched, and QIf and QQf are gain and phase matched, (IIf v,IQf), the output of the adder  182 , and (QQf,QIf), the output of the adder  184 , must be matched in phase. This is done by adding to the output of adder  182  a term Kri which is the output of adder 184 . This is explained simply by the equation: 
      cos( wt+PhiIR )=cos( wt )cos( PhiIR )+sin( wt ).sin( PhiIR ) 
     This means that to correct by a phase value PhiR, the imaginary part sin(wt) (addder  184  output) is taken and is multiplied by Kri=sin(PhiIR) and added to the adder  182  output cos(wt).cos(PhiIR). The operation of multiplication by cos(PhiIR) may be done before the adder  182  by changing Kii′=Kii.cos(PhiIR) and Kiq′=Kiq.cos(PhiIR) to save a multiplier. 
     Therefore, Ir=Kii.cos(PhIR).Iff.cos(wt−phi0+PhiIQ)+Kiq.cos(PhiIR).IQf.sin(wt−phi0)+Qr.Kri and Qr=Kqq.QQf.sin(wt−phi0−PhiQI)+Kqi.QIf.cos(wt−phi0) 
     When the received signal has a narrow bandwidth or a medium bandwidth (both of which may be termed for purposes of this description as narrowband, since they are narrow relative to the wideband signal previously discussed), like BW/2 rather than BW (for example a GSM cellular telephone signal), then two of the component branches of the receiver can be switched OFF while the other two branches are left ON with the digital Local Oscillator set to 0 or used to set gain control (AGC). (i.e, cos(wt)=1 or cos(wt)=AGC and sin(wt)=0). This then functions as a single Cartesian receiver. 
     The receiver of  FIG. 1  is shown in  FIG. 7  in such a mode. As shown in  FIG. 6 , the input to the receiver of  FIG. 7  is operated the same as that of  FIG. 1 , that is, the Local Oscillator  105 , and Local Oscillator quadrature generator  108  produce the same signals as earlier described, as do mixers  102  and  104  which serve to reduce the input channel to baseband and divide the spectrum into its I and Q quadrature components. In the receiver of  FIG. 6 , however, the IQ and QQ paths are disabled by multiplying the input signals thereto by 0. This allows only two paths to remain active, one path for each of the I and Q components of the complex, but narrow, or narrower, band input signal. 
     The active paths II and QI filter the input channel in analog low-pass filters  118  and  122 , respectively, perform analog to digital conversions in A/D converters  126  and  130  (which may be Sigma-Delta converters), and provide digital filtering in filters  134  and  138  to produce output signals lout and Qout which are then employed to derive the originally encoded information encoded by the transmitter. 
     As can be seen, in this manner both wideband and narrow band signals can be processed by the same receiver components and duplication of elements and data paths is avoided. 
     As previously noted, in this preferred embodiment a receiver is described in which the input signal is divided into four components (or two components in the case of medium or narrow bandwidth signals), but it is understood that a different number of components may be used depending upon the bandwidth of the input signal and the desired bandwidth of the components.