Abstract:
In transconductance amplifier arrangements used in CATV systems, it is often desirable to set the value of the output impedance to a value equal to the system impedance that is often 75Ω. prior art transconductance amplifiers often comprise an amplifier ( 6 ) with a current output using a feedback network ( 8 ) to set the gain value. An input of the feedback network ( 8 ) is coupled to the current output of the amplifier ( 6 ) and an output of the feedback network ( 8 ) is coupled to the input of the amplifier ( 6 ). In these prior art transconductance amplifier arrangements the output impedance decreases with increasing gain of the amplifier ( 6 ) used in the amplifier arrangement. This output impedance is normally very low (a few Ω or lower). By adding a further output current (i/N) to the output of the feedback network, it is obtained that the current through the feedback network ( 8 ) becomes dependent on the output current (i) of the amplifier. This dependence results in an increased output impedance.

Description:
BACKGROUND OF THE INVENTION 
     The present invention relates to an amplifier arrangement comprising an amplifier having an input terminal and a current output terminal for providing an output current in dependence on a signal at the input terminal, the amplifier further comprises a feedback path coupled between the current output terminal and the input terminal. 
     Such amplifier arrangement is known from the book “Design of High-Performance Negative-Feedback amplifiers” by Erst H. Nordholt, chapter 1, page 8, FIG. 4, Elsevier Publishing Company 1983, ISBN 0-444-42140-8. 
     Amplifier arrangements according to the preamble are used in several broadband applications, such as CATV systems. The prior art amplifier arrangement is intended for use with a signal source that behaves as a current source. The current forced by the input source into the amplifier, flows through the feedback path and causes a voltage across said feedback path. Due to the feedback arrangement, the current source in the amplifier supplies an output current which matches the sum of the current that flows in the feedback path and the current that flows into the load impedance of the amplifier arrangement. 
     The prior art amplifier arrangement shows good high frequency behavior. It further has a low output impedance, which even decreases with increasing gain of the amplifier. This low output impedance is not always desired. E.g. in CATV systems it is desired to drive the load impedance with an amplifier having an output impedance that matches the load impedance. In CATV systems this impedance is equal to 75Ω. This matching of load impedance and output impedance is desired to suppress reflections at the output of the amplifier of RF signals picked up by the coaxial cables in the CATV system. 
     An obvious solution to this problem could be connecting a resistor in series with the load impedance in order to obtain a desired value of the output impedance. This however would result in half the power generated by the amplifier being dissipated in said series resistor. 
     SUMMARY OF THE INVENTION 
     An object of the present invention is to provide an amplifier arrangement according to the preamble having an increased output impedance, without using series resistors dissipating a substantial fraction of the power generated by the amplifier. 
     To achieve the object of the present invention, the amplifier arrangement is characterized in that the amplifier comprises a further current output terminal for providing a further output current which is dependent on the current flowing through the current output terminal, and in that the second current output terminal is coupled to the output of the feedback network. 
     By introducing a further current output terminal, and coupling it to the output of the feedback path it is obtained that the voltage drop over the feedback path decreases with an increasing output current of the amplifier. This decrease of the voltage drop over the feedback path has as result that the output voltage of the amplifier arrangement decreases when the output current of the amplifier increases. This effect results in an increased output impedance of the amplifier arrangement. By using an amplifier arrangement according to the present invention is possible to obtain an amplifier arrangement of which the output impedance can be given a predetermined value without needing bulky transformers to obtain an impedance transformation. 
     An embodiment of the invention is characterized in that the amplifier comprises at least a first and a second transistor, in that a first main electrode of the first transistor being coupled to the current output, in that a first main electrode of the second transistor is coupled to the further current output, in that a second main electrode of the first and the second transistor are coupled to a bias source, and in that a control electrode of the first and second transistor are coupled to the input terminal of the amplifier. 
     The use of a parallel connected first and second transistor for realizing the current outputs results in a simple amplifier of which the ratio between the two output currents can be chosen by choosing a proper ratio of the emitter areas of the two transistors. 
     A further embodiment of the invention is characterized in that the amplifier comprises a third transistor, in that a first main electrode of the third transistor is coupled to a supply terminal, in that a second main terminal of the third transistor is coupled to the bias source, in that a control terminal of the third transistor is coupled to a further input of the amplifier, and in that a coupling element is coupled between the further input of the amplifier and a reference terminal. 
     By adding a third transistor, an amplifier with symmetric inputs is obtained. This makes it possible to use the amplifier arrangement with balanced input signals to convert said balanced signal into an unbalanced signal without needing transformers. 
     A further embodiment of the invention is characterized in that the feedback path and the coupling element comprise impedance elements with values having a ratio of N/(N+1), in which N is the ratio between the output current and the further output current. 
     In this embodiment the amplifier can be driven from two balanced current sources. 
     A still further embodiment of the invention is characterized in that the amplifier further comprises a voltage controlled current source with two input terminals and at least one output terminal, in that the first input terminal of the voltage controlled current sources is coupled to the control electrode of the first and second transistor, in that the second input terminal of the voltage controlled current source is coupled to the control electrode of the third transistor, and in that the output of the voltage controlled current source is also coupled to the control electrode of the third transistor. 
     In this embodiment the gain of the amplifier is increased due to a positive feedback introduced by the voltage controlled current source. Increasing the gain of the amplifier results in more accurate values of the transimpedance and the output impedance. An advantage of this particular way of increasing the gain of the amplifier is that the high frequency properties are hardly effected. However, care should be taken to ensure stability of the amplifier under all circumstances. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The invention will now be explained with reference to the drawings wherein same elements are indicated with same reference signs. 
     FIG. 1 shows a schematic of the principle of the amplifier arrangement according to the invention. 
     FIG. 2 shows a first embodiment of an amplifier arrangement according to the invention. 
     FIG. 3 shows a second embodiment of an amplifier arrangement according to the invention. 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     In the amplifier arrangement according to FIG. 1, an input current source  2  providing an input current i IN  is connected to an inverting input of an amplifier  6 . A non inverting input of the amplifier  6  is connected to a reference voltage source. 
     The current output terminal of the amplifier, carrying an output current i, is connected to the output of the amplifier arrangement to which a load impedance  10  with value R L  is connected. The current output terminal is further coupled to the input of the feedback path, which input is here a first terminal of a resistor  8  with value R 1 . The output of the feedback path, which output is here a second terminal of the resistor  8 , is connected to the inverting input of the amplifier  6 . 
     According to the inventive concept of the present invention, the further current output terminal, carrying a current i/N, is coupled to the output of the feedback path, which output is here the second terminal of the resistor  8 . 
     To determine the properties of the amplifier arrangement according to FIG. 1, the open voltage and the output impedance are determined. The open voltage can be determined by setting the value of the load resistor  10  to infinity. If the input impedance of the amplifier is regarded as very high, for the relation between the currents i IN  and i can be written:                  i   IN     +   i   +     i   N       =       0   ⇒   i     =       -     i   IN       ·     N     N   +   1                   (   1   )                                
     If the inverted input of the amplifier  6  is at virtual ground, for the output voltage V OUT  can be written:                V   OUT     =         i   ·     R   1       ⇒     V   OUT       =       -     i   IN       ·     N     N   +   1       ·     R   1                 (   2   )                                
     Consequently, the transimpedance of the circuit according to the invention is equal to N/(N+1) times the transimpedance of the prior art circuit whose transimpedance is equal to R 1 . 
     For determining the output impedance, first the short circuit current is determined. In a short circuit situation, the output voltage V OUT  is equal to zero. This means that no current can flow through resistor R 1  because the voltage across the resistor  8  is also equal to zero. Consequently, the current i/N is equal to −I IN . The short circuit current I k  is now equal to −N·I IN . For the output impedance Z OUT  of the amplifier arrangement according to FIG. 1 can be written:                Z   OUT     =         V   OPEN       i   K       =       R   1       N   +   1                 (   3   )                                
     From (3) can be seen that the output impedance strongly depends on N as the transimpedance only varies slightly with N. This makes it possible that the amplifier arrangement according to the invention can be given a suitable output impedance without having to use transformers, which are expensive and have unfavorable properties at very low and very high frequencies. 
     In the amplifier arrangement according to FIG. 2, the amplifier  6  is a differential amplifier constituted by transistors  13 ,  14  and  16 . The emitters of the transistors  13 ,  14  and  16  are connected to a bias current source  15  for receiving a predetermined bias current. The collector of the transistor  14  constitutes the first current output terminal of the amplifier carrying a current i 14 , and the collector of the transistor  13  constitutes the second current output terminal carrying a current i 13 . The emitter areas of the transistors  13  and  14  differ a factor N causing the collector current of the transistor  13  to be a factor N smaller than the collector current of the transistor  14 . 
     The amplifier arrangement according to FIG. 2 is designed for symmetrical input signals and a single ended output signal without using any transformers. The inductor  19  is present to allow the output voltage V OUT  to be larger than the supply voltage, which leads to an increased maximum voltage swing at the output. This is in particular important when the transistors  13 ,  14  and  15  are manufactured in a high-frequency process, which normally only allows low supply voltages. 
     For determining the output voltage of the amplifier arrangement according to FIG. 2, it is assumed that the gain of the amplifier  6  is large. Consequently, the voltage between the nodes X and Y is equal to zero. The small signal voltage at node Y will be equal to −i 2 ·R 2 . The voltage on the node X is then also equal to −i 2 ·R 2 . 
     If the amplifier according to FIG. 2 is unloaded, the small signal collector current of transistor  14  flows completely in resistor  8 . Applying Kirchhoffs law on the node X results into:                i   1     =       i   14     +       i   14     N               (   4   )                                
     From (4) the voltage on node Z can be calculated according to:                V   Z     =           -     i   2       ·     R   2       -       i   14     ·     R   1         =         -     i   2       ·     R   2       +       i   1     ·     N     N   +   1       ·     R   1                   (   5   )                                
     If R 2  is chosen equal to R 1 ·N/(N+1), (5) changes into                V   Z     =       (       i   1     -     i   2       )     ·     N     N   +   1       ·     R   1               (   6   )                                
     The short circuit current of the amplifier arrangement according to FIG. 2 can be calculated by setting the voltage on node Z equal to zero. The current in resistor R 1  will now be equal to V X /R 1 =−i 2 ·R 2 /R 1 . For the collector current of transistor  13  we have now:                i   13     =     -     (       i   1     -       i   2     ·     N     N   +   1           )               (   7   )                                
     For the collector current of transistor  14  can now be found:                i   14     =       -     (       i   1     -       i   2     ·     N     N   +   1           )       ·   N             (   8   )                                
     The short circuit current of the amplifier arrangement is equal to the sum of the current i 8  in the resistor  8  and the collector current of transistor  14 . Consequently the short circuit current is equal to:                      i   k     =       i   8     -     i   14                   =         -     i   2       ·     N     N   +   1         -     (         i   2     ·       N   2       N   +   1         -     N   ·     i   1         )                   =           -     i   2       ·     (       N   +     N   2         N   +   1       )       +     N   ·     i   1         =     N   ·     (       i   1     -     i   2       )                       (   9   )                                
     The output impedance Z OUT  of the amplifier arrangement can now easily be found by calculating the quotient of (6) and (9). By performing this calculation a value of R 1 /(N+1) is found. 
     The bias current of transistor  13  flows through the resistor  8 . This reduces the bias voltage at node X. However to maximize the output voltage swing of the amplifier arrangement, this bias voltage should be at about ½V DD . To avoid a reduction of the voltage swing, the bias current of transistor  13  should not flow through the resistor  8 , but should be supplied by some other means. It could be done by a current source, but this requires the availability of PNP transistors. These are however not always available in high frequency IC processes. The bias current of transistor  13  can also be provided by a resistor  18  with value R 3 , which is connected between the base of transistor  13  and the positive supply voltage. To eliminate the influence of the resistor  18  on the AC behavior of the amplifier arrangement an inductor could be connected in series with the resistor  18 . 
     Alternatively, the influence of the resistor  18  can be compensated by choosing the values of the resistors  8  and  22  according to:                R   2     =     1         N   +   1       N   ·     R   1         +     1     R   3                   (   10   )                                
     From the expressions derived above, design equations can be derived. For proper impedance matching the output impedance Z OUT  should be equal to the value R L  of the load resistance connected to the amplifier arrangement. Thus follows the first design equation 
     
       
           R   1 =( N+ 1)· R   L   (11) 
       
     
     Substituting (11) into (10) gives                R   2     =       1       1     N   ·     R   L         +     1     R   3           =         R   3     ·   N   ·     R   L           R   3     +     N   ·     R   L                     (   12   )                                
     Using (6) and the value of the output impedance of the amplifier arrangement calculated earlier, results into the output voltage V OUT  of the amplifier arrangement according to:                V   OUT     =         V   Z     ·       R   L         Z   OUT     +     R   L           =       (       i   1     -     i   2       )     ·       N   ·     R   L       2                 (   13   )                                
     Since the input currents are balanced, the input current is effectively doubled by (i 1 −i 2 ). This results in a value for the transimpedance gain G T  of the amplifier arrangement according to: 
     
       
           G   T   =N·R   L   (14) 
       
     
     The following procedure can now be followed to design the circuit. Since the value of the load impedance R L  with which the amplifier arrangement has to be used is known, the value of N can be determined from the required value of the transimpedance gain using (14). Next, the value R 1  follows from (11). Then the value R 3  is chosen such that the DC bias voltage at node X is about half the supply voltage. Finally the value R 2  follows from (12). The last two steps can be iterated a few times to achieve maximum output voltage swing. 
     FIG. 3 shown the embodiment of the invention according to FIG. 2, to which a voltage controlled current source  33  is added. The inputs of the voltage controlled current source  33  are connected to the nodes X and Y. The output of the voltage controlled current source is connected to node Y. 
     The voltage controlled current source adds a compensation voltage source between the nodes X and Y having a substantial equal amplitude but a different sign than the original voltage between the nodes X and Y. This causes the actual voltage between the nodes X and Y to decrease. The smaller voltage between the nodes X and Y reduces the distortion in the amplifier  12 . 
     As the use of the voltage controlled current source involves positive feedback, care should be taken to ensure stability of the amplifier. This can be done by limiting the transconductance of the voltage controlled current source to a predetermined value. 
     The voltage controlled current source  33  is realized by an emitter degenerated differential pair comprising transistors  36  and  38 . The emitters of the transistors  36  and  38  are connected to bias sources  30  and  32  and are connected to each other via a resistor  34  with value R 4 . The output of the voltage controlled current source which is constituted by the collector of the transistor  36  is connected to the node Y. The output current of the voltage controlled current source generates the compensating voltage in series with the amplifier inputs across the resistor  22  with value R 2 . 
     Simulations of the circuit showed an −3 dB bandwidth of 1.1 GHz, without showing any peaking. The bandwidth can be increased beyond 2 GHz by adding a small capacitor with value of ≈1 pF between node X and ground.