Abstract:
An electronic commutation motor including a stator unit configured as a pulse-modulation driven stator unit including at least two winding circuits in which an induced electromagnetic force is produced, and a phase switching switch; and a rotor unit powered by charge stored on a capacitor, wherein the capacitor is charged by diodes coupled between the capacitor the first winding circuits of the stator unit. In this way, the stator unit, serving as a first submachine, performs the function of power supply for the rotor unit serving as a second submachine, by charging of the capacitor via the diodes.

Description:
BACKGROUND OF INVENTION 
     1. Field of the Invention 
     This invention relates to a high-efficiency electric motor of electronic commutation type. 
     2. Discussion of the Background 
     High-efficiency electrical machines of electronic commutation type, hereinafter known as ECMs, operate with pulse modulation and generally at ultrasonic frequencies, with absorption of very high ripple current pulses. Without the use of a costly and bulky L-C filter in the feed line, the conducted and radiated electrical disturbance levels would be greater than allowed by current regulations. To reduce costs, the filter can be replaced by one of active type able to decouple the current absorbed by the electric motor from the battery current. A known and particularly effective implementation, in terms both of cost and performance, is to interpose between the battery and the ECM a step-up converter current-controlled by means of R FB  on the basis of control information C FB  originating from the ECM, which is compared with a velocity input V set  by known methods. This converter is characterised by operating with an output voltage V c  greater than the battery voltage V b  and by absorbing from the battery an essentially continuous current (of constant delivered power) with a ripple as small as desired, achieved by dimensioning the inductor L and the switching frequency by known methods. The waveform of the battery current i b  is shown in FIG. 2, which shows the typical times associated with the operation: 1/T is the switching frequency, T on  and T off  are the on and off times of the electronic power switch P (FIG. 1); also shown are the ripple superimposed on the mean absorbed current and the composition of i b , consisting of the sum of i P  and i D , this latter being integrated by the capacitor C to provide a mean current i 2  (from an essentially continuous V c ) which powers the ECM. 
     SUMMARY OF THE INVENTION 
     The object of the invention is to achieve the operability of the schematic of FIG. 1 essentially in terms of the waveform of the current absorbed from the battery, while significantly reducing cost and bulk by eliminating the inductance L and the switch P. As it is not possible to eliminate these components from an operational viewpoint, the invention proposes a solution which utilises certain switches and certain windings of the ECM, already present for its normal operation, to also perform the function of switch P and inductance L. 
     This object is attained according to the invention by a high-efficiency electric motor of electronic commutation type, having a single stator unit and a single rotor unit, including a first electrical submachine and a second electrical submachine, in which: 
     the first submachine is fed directly by a voltage source and is associated with a sensor for measuring the current absorbed from the feed; said first submachine including at least two windings characterised by an inductance, a resistance, an induced electromotive force and a switch connected in series; 
     the second electrical submachine is fed uniquely by a capacitor which is charged at a controlled voltage; 
     for each of said first windings there is provided a diode, having one of its poles connected to the end of the respective winding, which is connected to said switch, and the remaining pole connected to one of the ends of the capacitor thus charged at a controlled voltage; 
     the first submachine is pulse-modulation driven to obtain a closely DC current absorption from said voltage source with harmonics content as low as desired and, by charging the capacitor at the voltage via the diodes, said first submachine provides the unique power supply for said second submachine. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     A more complete appreciation of the invention and many of the attendant advantages thereof will be readily obtained as the same becomes better understood by reference to the following detailed description when considered in connection with the accompanying drawings, wherein: 
     FIG. 1 is a schematic block diagram of an electric motor of known type; 
     FIG. 2 shows the waveforms of the current through the motor of FIG. 1; 
     FIG. 3 is a schematic block diagram of the electric motor according to the invention; 
     FIGS. 4 and 5 are illustrations of two electromagnetic structures forming the electric motor of the invention; 
     FIG. 6 is a schematic block diagram of a specific electric motor of the invention; 
     FIG. 7 is an illustration of waveform diagrams illustrating the phase emfs of the motor of the invention; 
     FIG. 8 is a simplified schematic block diagram corresponding to that of FIG. 6; 
     FIG. 9 is a further schematic block diagram of the machine of the invention; 
     FIG. 10 is a waveform diagram; 
     FIG. 11 is a schematic block diagram of the motor of the invention provided with protection devices; 
     FIGS. 12 and 13 are further waveform diagrams; and 
     FIG. 14 is a schematic block diagram of an additional circuit for the electric motor of this invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Referring now to the drawings, wherein like reference numerals designate identical or corresponding parts throughout the several views. 
     Essentially according to the invention, the inductance L and switch P of FIG. 1 are integrated into a suitably structured ECM, controlled and dimensioned to add to its electric motor function the function of active filter, so covering by itself the overall operability of the schematic of FIG. 1. The first feature of the ECM proposed by the invention (FIG. 3) is that it operates as two submachines which mechanically combine their contributions at the same rotor of the ECM whereas electrically they operate and are controlled as two separate machines. The first, known hereinafter as M1, is powered by the battery at voltage V b , whereas the second, known hereinafter as M2, is powered by a capacitor C charged to a voltage C c  by the operation of M1 as described hereinafter. The scheme is completed by the fast diodes D connected to the capacitor C as in FIG. 3. The velocity input V set  and the signals of the Hall position sensors are also shown. 
     The second feature is that in order to also perform the function of the inductance L and the switch P of FIG. 1, the submachine M1 must be designed with a unipolar structure with two or more windings (depending on the number of phases to be determined and the number of windings to be powered in parallel) with the magnetic coupling between them as loose as possible. The inductances of its windings and the switches P already proposed for their normal PWM driving provide the L and P functions of FIG. 1. 
     The third feature is that the submachine M2 can have a different number of phases and windings than the submachine M1, with any magnetic coupling between them, but magnetically decoupled from the windings of M1. 
     The fourth feature is that the driver of M2 is totally independent of that of M1. It can therefore be of unipolar, bridging, linear or PWM type and is characterised by having a control function (for example a control feed-back on V c ) which ensures that under all operating conditions the current induced by the operation of M1 via the diodes D is totally absorbed by M2. Without limiting the generality of the aforedescribed principle of operation, for greater clarification and for providing the main design principles, reference will be made to a two-phase battery powered unipolar brushless motor of permanent magnet type. 
     Two electromagnetic structures which implement the aforesaid magnetic coupling conditions are shown in FIGS. 4 and 5 by way of non-limiting example. 
     In particular, for the same nominal ECM operating conditions and the same number of poles, the structure of FIG. 5 has a lower phase inductance and a lesser demagnetising reaction (1/3 of that of the structure of FIG. 4). 
     The specific schematic which achieves the said principles (FIG. 1 and FIG. 3) is shown in FIG. 6. To complete the control electronics, in addition to that already described it includes two signals V m2  for operating by known circuits (clamping circuits) a protection at overvoltages exceeding the VDSS allowed by the switches P2. These latter together with other circuit details are known and do not form part of the inventive idea, and will therefore not be referred to hereinafter. The chosen two-phase structure is for example of known type with four unipolar windings powered as two single-phase machines (at full half-wave). The first single-phase machine (consisting of PHASE 1 and PHASE 3) covers the role of the submachine N2 and is powered at V c . The emfs of each phase (e F1 , e F2 , e F3 , e F4 ) are shown in FIG. 7, where it can be seen that they are out of phase by 90 electrical degrees. 
     The magnetic structure, the seat of the magnetic flux generated by the currents in each winding of the submachine M1 (identified in FIGS. 4, 5 and 6 as PHASE 1 and PHASE 3), must be such as to ensure that the inductances of these windings are as mutually decoupled as possible to prevent absorbed current gaps during switching between one winding and the next in the driving sequence (a known problem when mutual inductance exists between the two) and that the inductive couplings with the sindings of M2 are marginal. This is achieved by the presence of non-wound decoupler teeth (indicated by T d ) and winding the two phases (PHASE 1 and PHASE 3) on physically separate teeth (see FIGS. 4 and 5). Said M2 windings also operate as an electric motor generating an active torque, as they suitably engage the pertinent emf half-wave by known methods (e.g. suitable decoding of Hall position sensors). The magnetic structure, the seat of flux generated by the currents in each winding of the submachine M2 (identified in FIGS. 4, 5 and 6 as PHASE 2 and PHASE 4), must ensure in this case a very tight magnetic coupling between them to enable the stored magnetic energy (from the windings which cease to conduct to those which begin to conduct) to be transferred during switching with minimum losses via the diodes D2 (known operation). This is achieved by winding said phases on the same teeth (see FIGS. 4 and 5). 
     As the two submachines operate in parallel in providing the desired mechanical power it is generally advantageous to dimension them such that, at least under nominal conditions, both the mechanical power supplied and the losses are divided into equal parts. 
     The design data for said operating point (n) are: 
     P mach (n) mechanical power 
     RPM(n) velocity 
     η(n) efficiency 
     V b  feed voltage 
     Knowing the design data, the geometry and the materials chosen for constructing the machine, the iron, ventilation and friction losses P fe ,v,a(n) can be predicted by known methods. 
     The value of R FB  is chosen such that the voltage drop across it can be considered negligible as a first approximation, so that to simplify the calculations the diode is simulated as an ideal diode with a resistor equal to R pi  in series (see FIG. 8). 
     By way of example, for the machine of FIG. 8 the equivalent scheme shown in FIG. 6 can be used, which shows the essential components for dimensioning the two machines, these being: 
     L f1  inductance of each winding of M1 
     R f1  resistance of each winding of M1 
     E f1  mean emf per half wave at the nominal velocity of each winding of M1 
     R P1  internal resistance of the power switch (e.g. MOSFET) for each winding of M1. 
     FIG. 8 also shows the corresponding elements for M2. 
     The two submachines (M1) and (M2) must be designed as follows. Dimensioning of submachine M1: 
     The first element immediately obtainable is i 1 (n) from 
     
         η(n)=P.sub.mech(n) /V.sub.b(n) i.sub.1(n) 
    
     hence 
     
         i.sub.1(n) =P.sub.mech(n) /V.sub.b(n) N.sub.(n)            (eq. 1) 
    
     Equation 1, together with cost considerations and other known operational aspects of the switch P1, enables its type to be identified and hence R P1  to be qualified as an item of data. Having identified i 1 (n) and R P1 , E f1  (n), E f1 (1000) and R f1 (n) can be obtained. From the known relationship P gap  =P mech  +P fe ,v,a =E·I and remembering that the power has to be distributed equally between the machines M1 and M2, the for M1: 
     
         E.sub.f1(n) i.sub.1(n) =[V.sub.b(n) i.sub.1(n) η.sub.(n) +P.sub.fe,v,a(n) ]/2 
    
     hence 
     
         E.sub.f1(n) =1/2[V.sub.b(n) η(n)+P.sub.fe,v,a(n) /i.sub.1(n) ] 
    
     which by replacing i 1 (n) by eq. 1 gives: 
     
         E.sub.f1(n) =1/2V.sub.b(n) η(n)[1+P.sub.fe,v,a(n) /P.sub.mech(n) ](eq. 2a) 
    
     As P fe ,v,a(n) is negligible compared with P mech (n), (eq. 2a) can 5 be rewritten as 
     
         E.sub.f1 (n)≈1/2V.sub.b(n) η(n)                (eq. 2b) 
    
     from which E f1 (1000) can be obtained as follows: 
     
         E.sub.f1 (1000)=[E.sub.f1(n) /RPM(n)]·1000        (eq. 3) 
    
     Hence using known formulas the number of turns of the winding and the value of L f1  can be calculated. To obtain R f1 (n) an energy balance can be used in which the machine M1 absorbs 50% of the total power. Hence: 
     
         [E.sub.f1(n) +(R.sub.f1 +R.sub.P1)i.sub.1(n) ]i.sub.1(n) =1/2V.sub.b(n) i.sub.1(n) 
    
     giving 
     
         E.sub.f1 (n)+(R.sub.f1 +R.sub.P1)i.sub.1(n) =1/2V.sub.b(n) 
    
     from which 
     
         R.sub.f1 =[V.sub.b(n) /2-E.sub.f1(n) ]/i.sub.1(n) -R.sub.P1 (eq. 4) 
    
     Dimensioning of submachine M2: 
     Defining T on  and T off  as the on and off times of the switches P1 respectively, 
     
         T=T.sub.on +T.sub.off, D=T.sub.on /T, T.sub.off /T=(1-D). 
    
     Regardless of the voltage V c  across the capacitor C, its charging current can be obtained from the always valid relationship: 
     
         i.sub.2 =i.sub.1 T.sub.off /T=i.sub.1 (1-D) 
    
     which at the nominal operating point can be written as 
     
         i.sub.2 (n)=i.sub.1(n) (1-D(n))                            (eq. 5) 
    
     The relationships between M2 and M1 for their respective characterising elements can now be obtained. Remembering the condition of equal power, then: 
     
         E.sub.f2(n) ·i.sub.2(n) =E.sub.f1(n) ·i.sub.1(n) 
    
     hence 
     
         E.sub.f2(n) =E.sub.f1(n) ·i.sub.1(n) /i.sub.2 (n) 
    
     and finally 
     
         E.sub.f2(n) =E.sub.f1(n) /(1-D.sub.(n))                    (eq. 6) 
    
     Remembering also the condition of equal dissipated power, then: 
     
         R.sub.f2(n) ·i.sub.2 (n).sup.2 =R.sub.f1(n) ·i.sub.1(n).sup.2 
    
     hence 
     
         R.sub.f2(n) =r.sub.f1(n) ·(i.sub.1(n) /i.sub.2(n)).sup.2 
    
     and finally 
     
         R.sub.f2(n) =R.sub.f1(n) /(1-D.sub.(n)).sup.2              (eq. 7) 
    
     The only unknown is D.sub.(n), which can be obtained from 
     
         Δi.sub.1,Ton =Δi.sub.1,Toff 
    
     from which, having assumed RD=Rp1 
     
         (V.sub.b -[E.sub.f1 +(R.sub.f1 +R.sub.P1)i.sub.1 ])T.sub.on /L.sub.f1 T=(R.sub.P1i1 +V.sub.c -[V.sub.b -(E.sub.f1 +R.sub.f1 i.sub.1)])T.sub.off /L.sub.f1 T                                               (eq. 8) 
    
     Putting A=V b  -[E f1  +(R f1  +R P1 )i 1  ], then: 
     
         D=(V.sub.c -A)/V.sub.c                                     (eq. 9) 
    
     Hence, remembering (eq. 4), 
     
         1-D.sub.(n) =v.sub.b(n) ·V.sub.c(n) /2            (eq. 10) 
    
     from which it can be seen that having fixed V b , (1-D.sub.(n)) is defined unabmiguously by V c (n). 
     The three ensuing conditions help to define V c (n) unambiguously. These are: 
     Condition 1 
     In order for current not to circulate through that winding of the submachine M1 which with its emf, the sum of the motional part E f1 (n) and the transformer part E m1 (n) due to undesirable coupling between the windings of the submachine M1 and between these and those of the submachine M2, would give a negative contribution to the development of mechanical power, the voltage V DS1 (off) across the power switch P 1 (off) connected to said winding must be less than the voltage across the capacitor C. Only in this manner can the diode D 1 (off) be polarised inversely and hence current cannot pass therethrough. The following condition must therefore be satisfied (see FIGS. 8 and 9): 
     
         V.sub.c(n) ≧V.sub.DS1(off) =V.sub.b(n) +E.sub.f1(n) +E.sub.m1(n)(cond. 1) 
    
     Condition 2 
     As the maximum voltage v DS2 (off) across the power switch P2 occurs during the time interval in which that winding of the submachine M2 connected to it is inactive, then: 
     
         V.sub.DS2(off) =V.sub.c(max) +E.sub.f2(max) =2V.sub.c(max) ; 
    
     hence in order for the rupture voltage V DSS2  of the power switch P2 not to be exceeded, the following condition must be satisfied: 
     
         2V.sub.c(max) &lt;V.sub.DSS2                                  (cond. 2) 
    
     Condition 3 
     Remembering that: 
     the coupling between the windings of the submachine M2 must, as stated, be as high as possible; 
     the transfer of magnetic energy, which occurs through D2 during switching between the windings of submachine M2, is less dissipative the higher the difference between the feed voltage, which in this case is V c , and the transient overvoltage V ts2 (off,t) (made as close as possible to V DSS2  by said clamping circuits) which appears across the power switch P2 when it opens; 
     the cost of the capacitor C increases with its rated voltage; it is apparent that V c (n) must be as low as possible (cond. 3). 
     Given that in practice: 
     
         E.sub.f1(n) +E.sub.m1(n) ≈1/2V.sub.b(n) 
    
     then (see (cond. 1)) 
     
         V.sub.c(n) ≈3/2 V.sub.b(n)                         (eq. 11). 
    
     From (eq. 10) and (eq. 11) the following are also obtained: 
     
         i.sub.2(n) =i.sub.1(n) /3                                  (eq. 12.1) 
    
     
         E.sub.f2(n) =3E.sub.f1(n)                                  (eq. 12.2) 
    
     
         R.sub.f2(n) =3.sup.2 R.sub.f1(n)                           (eq. 12.3) 
    
     
         P.sub.p2(n) =3.sup.2 r.sub.P1(n)                           (eq. 12.4) 
    
     Equations 12.1-12.4, which unambiguously determine the dimensioning of the submachine M2, show an interesting aspect from the constructional viewpoint, namely that for the two submachines, wire of the same cross-section can be used, with a different number of wires in parallel for the two submachines. 
     If 1 m  is the mean turn length identical for all windings of the two submachines, S c1  the wire cross-section of each winding of the submachine M1 and S c2  the wire cross-section of each winding of the submachine M2, then: 
     
         R.sub.f1 =ρ(l.sub.m N.sub.s1)/C.sub.c1                 (eq. 13.1) 
    
     
         R.sub.f2 =ρ(l.sub.m N.sub.s2)/C.sub.c2                 (eq. 13.2) 
    
     Given that from (eq. 12.2) it can be deduced that the number of turns N s1  of each winding of the submachine M1 must be 1/3 the number N s2  of each winding of the submachine M2: 
     
         N.sub.s1 =1/3·N.sub.s2                            (eq. 13.3) 
    
     From (eq. 12.3) and (eq. 13.1-13.3): 
     
         ρ(l.sub.m ·N.sub.s2)/S.sub.c2 =3.sup.2 ρ(l.sub.m ·N.sub.s1)/S.sub.c1 =3.sup.2 ρ(l.sub.m ·N.sub.s2 /3)/S.sub.c1 
    
     hence 
     
         S.sub.c1 =3S.sub.c3                                        (eq. 14) 
    
     This latter shows that the winding of the submachine M1 can be formed by positioning in parallel three wires of cross-section identical to that of the single wire used for the winding of the submachine M1. A PWM control strategy at fixed frequency is normally implemented on step-up converters of the type shown in FIG. 1. Given that, as clarified in the description of the inventive idea, the function of the inductor L of FIG. 1 is performed by windings which are the seat of induced emf, a strategy such as the aforegoing would make it difficult to contain the battery current ripple within predetermined limits. For this reasom the control strategy adopted is of hysteresis type, which acts only on the on phase of the submachine M1 and, in accordance with known methods, maintains the current is absorbed by the ECM, as measured through the resistor R FB , within predetermined maximum and minimum values such as to make the ripple as small as desired compatible with the technical limitations related to the state of the art of the switching devices used. This naturally means that the switching frequency of the power switches of the submachine M1 is not set but is directly related to its electrical parameters (inductance, emf, feed voltage). Conveniently, a control strategy is used for the voltage V c  across the capacitor C which for each delivered torque and rotational velocity condition satisfies the said (cond. 1), while maintaining the difference between V c  and V DS1 (off) as small as desired by known methods. The said strategy enables the battery current to be fully-controlled during switching between windings of the submachine M1. If during switching between windings of the submachine M1 it happens that the current in the phase which is switched off decreases more rapidly than the current increase in the phase which is switched on, the current is fails to below the minimum set value. If in contrast when one phase is switched off the current decreases more slowly than the current increase in the phase which is switched on, the is control maintains it within the preset limits. To obtain this condition it is necessary that during the switching time the average value of E f1 , known as E f1 ,avg is such that 
     
         V.sub.b -E.sub.f1,avg &gt;V.sub.c -(V.sub.b -E.sub.f1,avg) 
    
     As V c  ≈3/2 V b , necessarily E f1 ,avg &lt;0.25 V b . 
     Given that this is achieved by simply anticipating switching (already necessary for operation of the submachine M2 and easily implemented), the absorbed current ripple is hence easily controllable in any event. A filter for eliminating conducted and radiated electrical disturbances is conveniently positioned in the ECM feed line (see FIG. 11) and is of much smaller cost and size than that required for an ECM which does not implement the inventive idea. The simplest way of protecting a battery-powered ECM is to connect a power diode in series with the operating relay. Besides being costly and bulky, this diode introduces a voltage drop (typically 0.7 Volt) and hence reduces the EM efficiency (for equal absorbed power). The operating relay, which is key-operated, has to withstand a switch-on current which is so high as to require: unacceptable overdimensioning. According to the schematic shown in FIG. 11 the ECM is instead directly powered by the battery via the relay RL controlled by the electronic control unit ECU. A lower-power diode D P  and a ballast resistor R z  are connected as shown in FIG. 11. Given that the electronic control unit which controls the relay RL is key-powered via D p , the ECM is protected against polarity inversion. The ballast resistor R z  prolongs the duration of the current pulse which charges the capacitors C and C F  when the starting switch is operated, so limiting the extent of the dV/dt to which the capacitors are subjected and preventing passage of destructive current through the switch. The electronic control unit ECU measures the voltage across the resistor R z  and enables the relay RL only when this voltage, and hence the switch-on current, fails below a predetermined safety level. 
     Referring to eq. (11) V c  ≈3/2 V b , there are some cases (for instance to lower the rms current through the capacitor C, to lower the current through the switches of submachine M2, etc.) in which it is necessary to have V c  &gt;3/2 V b . In that case it could happen that during the commutation between the phases of the submachine M1, the current in the phase which is switched off decreases more rapidly than the current increase in the phase which is switched on: the battery current will fall out of the prescribed tolerance-band. 
     To avoid the fall of the battery current it is necessary to add an electronic circuit (FIG. 14) to control the current in the phase which is switched off. This is attained, as described below, by artificially prolonging the conduction interval of each phase of submachine M1, feeding to the gate of the corresponding MOSFET a clock signal logically anded with the pwm signal that normally controls the phases of submachine M1 in order to maintain the battery current within the prescribed tollerance-band. The decrease of the phase current vs time (slope) is controlled at a value such to avoid battery current to fall out of the above mentioned. 
     The logic keeps the MOSFET definitively off when the phase current reaches zero. The behaviour of the circuit will be explained for one of the two phases (named 1) of submachine M1, providing and complementary circuitry is used for the other(s). 
     Referring to FIGS. 12 and 13, let the phase 1 switched off. 
     V D1  =voltage at the drain of MFT1 
     V C  =voltage across the capacitor C 
     clock=square wave with duty-cycle value less than 50% the duty-cycle of the pwm signal and frequency value at least greater than three times the frequency of pwm signal 
     hall=the Hall effect sensor signal which switches on phase 1 
     pwm=signal which normally controls the phases of submachine M1 in order to maintain the battery current within the prescribed tolerance-band. 
     When HALL goes down to &lt;low&gt;, MFT1 is switched (momentary) off, V D1  becomes greater than V C , b 1  goes to &lt;high&gt;, y i  goes to &lt;high&gt; and q 1  will latch clock, out 1  =clock:MFT1 will be controlled by pwm anded with the clock one (see FIG. 12). 
     When the current through the phase 1 reaches zero and q 1  (latched to clock) switches off MFT1, V D1  cannot override V c , B 1  goes to &lt;low&gt; and when clock goes to &lt;low&gt;, y 1  goes to &lt;low&gt;; suddenly q 1  will go to &lt;low&gt;, out 1  will go down to &lt;low&gt; and MFT1 will be definitively switched off (see FIG. 13).