Abstract:
A waveguide-to-microstrip transition package (30) for processing electromagnetic wave signals includes a waveguide (32) for directing the signals to the input of the waveguide (32). A substrate (34) overlaps the input of the waveguide (32) to form a hermetic seal. A metallized probe (36) conducts the signals to a microstrip line (40) and is patterned upon the substrate (34). The transition (30) also includes an iris (48) formed from a metallized pattern on the opposite side of the substrate (34) from the probe (36). The special design of the probe (36), the structure of the iris (48) and the wave guide cavity (46) above the probe (36) allow impedance matching and efficient signal transfer from waveguide (32) to microstrip line (40) or from microstrip line (40) to waveguide (32).

Description:
This invention was made with government support under grant DAAL01-95-C-3536 awarded from the DARPA/MAFET program. The government has certain rights in the invention. 
    
    
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention is generally related to monolithic microwave/millimeter waveguide devices and more particularly to packaging waveguide-to-microstrip transitions for microwave/millimeter waveguide devices. 
     2. Discussion 
     In the past, several waveguide-to-microstrip design methodologies have been proposed in an effort to introduce an efficient transition from waveguide to microstrip. The need for such a transition is prompted by the numerous applications it has in present mm-wave (mmW) and microwave/millimeter wave integrated circuit (MMIC) technologies. The increased use of low-cost MMIC components such as low-noise and power amplifiers, in both military and commercial systems continues to drive the search for more affordable and package-integrable transitions. 
     The current method of signal reception and power transmission within the mmW system is the rectangular waveguide which has a relatively low insertion loss and high power handling capability. In order to keep the overall package cost to a minimum, there is a need for a transition which is mechanically simple and easily integrated into the housing while maintaining an acceptable level of performance. 
     Current designs have used transitions which were based on stepped ridged waveguides as discussed, for example, in: S. S. Moochalla and C. An, &#34;Ridge Waveguide Used in Microstrip Transition&#34;, Microwaves and RF, March 1984; and W. Menzel and A. Klaassen, &#34;On the Transition from Ridged Waveguide to Microstrip&#34;, Proc. 19th European Microwave Conf., pp. 1265-1269, 1989. Other designs used antipodal finlines which were discussed, for example, in: L. J. Lavedan, &#34;Design of Waveguide-to-Microstrip Transitions Specially Suited to Millimeter-Wave Applications&#34;, Electronic Letters, vol. 13, No. 20, pp. 604-605, September 1997. 
     Moreover, current designs have used probe coupling which was discussed, for example, in: T. Q. Ho and Y. Shih, &#34;Spectral-Domain Analysis of E-Plane Waveguide to Microstrip Transitions&#34;, IEEE Trans. Microwave Theory and Tech., vol. 37, pp. 388-392, February 1989; and D. I. Stones, &#34;Analysis of a Novel Microstrip-to-Waveguide Transition/Combiner&#34;, IEEE MTT-S Int&#39;l Symposium Digest, San Diego, Calif., vol. 1, pp. 217-220, 1994. 
     These current designs suffer from such disadvantages as varying degrees of mechanical complexity. Some of the current transitions are bulky and use several independent pieces that must be assembled in various steps. Additionally, they may require more than one substrate material with multilevel conductors and high-tolerance machining of background housing components such as waveguide steps/tapers, or precise positioning of a backshort. Such precise positioning requirements produce extensive bench tuning after fabrication. Also, current designs require a separate waveguide window and several hermetic sealing process steps to achieve hermetic sealing of the component. These disadvantages render current designs expensive and difficult to integrate into the package. 
     SUMMARY OF THE INVENTION 
     A waveguide-to-microstrip transition for processing electromagnetic wave signals includes a waveguide for directing the signals to a waveguide input. A substrate covers the waveguide input and is hermetically sealed to the waveguide. A probe on the substrate overlies the waveguide input. 
     In another embodiment, the waveguide-to-microstrip transition includes an iris connected to the substrate for substantially matching the impedance between the probe and a microstrip line. 
     Additional advantages and features of the present invention will become apparent from the subsequent description and the appended claims, taken in conjunction with the accompanying drawings in which: 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a diagrammatic perspective of the waveguide-to-microstrip transition; 
     FIG. 2 is a diagrammatic perspective of the waveguide-to-microstrip transition wherein the internal portions of the package are revealed; 
     FIG. 3 is an exploded perspective view of the waveguide-to-microstrip transition of the present invention; 
     FIG. 4A is a top view of the waveguide-to-microstrip transition showing the network topology; 
     FIG. 4B is a side view of the waveguide-to-microstrip transition depicting the waveguide and cavity dimensions; 
     FIG. 5 is a Smith chart used to determine the W-band dimensions for the iris; 
     FIG. 6 is an X-Y graph illustrating the predicted results of the Q-band transition; 
     FIG. 7 is an X-Y graph showing the measured data of two back-to-back Q-band transitions; 
     FIG. 8 is an X-Y graph showing the predicted results of the W-band transition; and 
     FIG. 9 is an X-Y graph showing the measured data of two back-to-back W-band transitions. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Referring to FIG. 1, a waveguide-to-microstrip transition package is generally shown at 30. The opening of waveguide 32 allows electromagnetic millimeter/microwave signals to reach substrate 34. A probe 36 is etched onto the top of substrate 34. Probe 36 terminates with a first stub 38. Transition 39 indicates where probe 36 transitions into a microstrip line 40. Microstrip line 40 has a second stub 42 and a third stub 44; both stubs can be either an open or a shorted element. Above substrate 34 is a cavity 46, and below substrate 34 is an iris 48. 
     FIG. 2 shows the package 30 with its internal structure revealed. A ring frame 50 which is placed on top of base 52 defines cavity 46. Probe 36 which is etched on the backside of substrate 34 eliminates the need for separate assembly steps for the substrate-to-probe adhesion. The etching can be done by a photolithographic or other such process known in the art. Substrate 34 is self-aligning as indicated at location 54 which is advantageous particularly for applications requiring tight tolerances such as W-band packaging applications. 
     Substrate 34 overlaps waveguide input 63 which makes a natural hermetic seal as indicated at location 56. Iris 48 on waveguide input 63 provides matching between probe 36 and waveguide input 63 as shown at location 58. In addition, iris 48 allows the formation of a cavity 46 above the probe 36, resulting in the backshort length to be a less critical dimension. Location 59 depicts the elimination of glass-to-metal seal contact to substrate. 
     Referring to FIG. 3, package 30 is constructed in three parts which has the decided advantage of a lower assembly cost. A cover 60 is placed upon ring frame 50. Cover 60 provides the covering for both the RF components of package 30 as well as for the backshort for transition 39. An opening 61 is provided for the waveguide. Moreover, a trough 62 allows substrate 34 to be accurately aligned with base 52. Substrate 34 is eutectically soldered or epoxied to base 52 for a hermetic seal. A second substrate 64 with the same configuration as substrate 34 is shown. 
     Optimal coupling of RF power to and from package 30 is accomplished by making use of available iris resonances due to excited higher-order modes and the terminating of the microstrip line 40 in a short circuit at the edge of iris 48 (of FIG. 2) using first stub 38. Thus, the need for high-tolerance backshort positioning is obviated. Impedance matching to the microstrip port 69 is accomplished using microstrip elements designated with reference numerals 40, 42, and 44, rendering a very low-profile design. In this context, a very low-profile design indicates a planar microstrip design versus other designs such as ridged waveguide, or waveguides/coaxial/microstrip transitions. 
     Ring frame 50 encloses transition 39 with the exception of the opening for the microstrip line 40. Ring frame 50 which provides the perimeter for cavity 46 is assembled along with substrate 34 in one step. Another feature of transition 39 is that cover 60 is an integral part of package 30, and can be laser-welded in place, thus making transition 39 a fully integrated part of package 30 requiring no special assembly steps. These features render transition 39 to be very low-cost and readily integrable into typical microwave and mmW multi-chip assembly (MCA) packages. 
     In the preferred embodiment: substrate 34 is composed of alumina; with etched gold probe 36 and etched gold iris 48; ring frame 50 is a composition of Alloy 48 and 46; base 52 is of composition of AlSiC (cast) and CuMo (stamped) corresponding respectively. However, it is to be understood that the present invention is not limited to only those compositions referenced above, but includes other materials which produce similar results. For example, substrate 34 may also have the following compositions (but is not limited to): fused silica, duriod, or z-cut quartz. 
     Referring to FIG. 4A, probe 36 is situated along the E-plane of the waveguide, and is terminated in a short structure (i.e., first stub 38) coincident with edge 66 of iris 48 and connects to the main microstrip line (not shown). This ensures a zero voltage condition at edge 66, and in turn, maximum voltage across the opening of iris 48 and RF coupling to the signal transmitting line. Preferably, first stub 38 is a ninety degree stub. The probe 36, the stubs (38, 40, 42) and iris (48) are patterns formed from etching of gold metallization of both sides of the substrate 34. 
     The choice of iris height 67 (H iris ) and iris width 68 (W iris ) determines the upper bound for the bandwidth of the transition. Iris 48 was modeled as a shunt circuit, where the equivalent circuit parameters model the storage of susceptive energy caused by the non-propagating higher-order modes excited at the discontinuity. These shunt parameters are determined using a variational method such as that described in R. E. Collin, Field Theory of Guided Waves, McGraw-Hill, New York, ch. 8, 1960. Because of this total admittance, iris 48 has resonances of its own which can in turn be used to broaden the bandwidth of the transition (see, L. Hyvonen and A. Hujanen, &#34;A Compact MMIC-Compatible Microstrip to Waveguide Transition&#34;, IEEE MTT-S Int&#39;l Symposium Digest, San Francisco, Calif., vol. 2, pp. 875-878, 1996. 
     The optimal choice of dimensions of iris 48 is accomplished using a 3D electromagnetic simulator based on Finite Element Method (FEM), such as Ansoft&#39;s Maxwell Eminence or Hewlett-Packard&#39;s HFSS. 
     Matching of the impedance presented by iris 48 to the microstrip port 69 is accomplished by using two symmetrical shunt lines 72 and 74 which are short-circuited using second and third stubs (42 and 44). Shunt lines 72 and 74 are a predetermined distance 70 (L 1 ) away from edge 65. This distance is chosen so that at point &#34;a&#34;: 
     
         Y.sub.a =Y.sub.0 +jB.sub.a                                 (EQ1) 
    
     where Y 0  is the characteristic admittance of the microstrip line 40. The lengths of shunt lines 72 and 74 (L 2 ) are chosen such that they each present: ##EQU1## to microstrip line 40 at f 0 , where B a  is the susceptance from (EQ 1). The use of two symmetrical shunt lines 72 and 74 in parallel assist in keeping the response broadband due to the higher series reactance seen by microstrip line 40: ##EQU2## In alternate embodiments, fine tuning of the response with respect to f 0  is implemented by varying W iris  68 accordingly. 
     Referring to FIG. 5, the input impedance referenced to the near edge of the iris is plotted on a Smith Chart parametrically as a family of curves for each H iris  as a function of W iris , Z in  (W iris )| H   iris . For the W-band design, choosing a curve with the least variation in Z in  (W iris ) H   iris  is equivalent to choosing the iris dimensions that will afford the broadest bandwidth for the matched transition. 
     Curve 100 depicts the following three points which pair H iris  with W iris  : (20.0 mils, 70 mils); (20.0 mils, 80 mils); and (20.0 mils, 90 mils). Curve 102 depicts the following three points which pair H iris  with W iris  : (25.0 mils, 70 mils); (25.0 mils, 80 mils); and (25.0 mils, 90 mils). Curve 104 depicts the following three points which pair H iris  with W iris  : (27.5 mils, 70 mils); (27.5 mils, 80 mils); and (27.5 mils, 90 mils). Curve 106 depicts the following three points which pair H iris  with W iris  : (30.0 mils, 70 mils); (30.0 mils, 80 mils); and (30.0 mils, 90 mils). Curve 106 exhibits at H iris  equal to 30.0 mils the least variation as a function of W iris . When the iris is implemented with an H iris  of 30.0 mils and an W iris  of 80 mils, the present invention provides for broadband performance. 
     Referring to FIG. 4B, the dimensions of cavity 46 (i.e., cavity height 78 and cavity width 80) are selected such that its modal resonances are not too close to the operating frequency. Usually, resonances are chosen such that: ##EQU3## where f 0  is the center operating frequency, and the f res   i  are the two closest resonances bounding the center frequency. Because of the relative isolation of cavity 46 from waveguide 32 due to iris 48, the present invention has the distinct advantage that the exact height of the backshort (i.e. H C  78) is not crucial to the electrical performance of the transition. 
     A Q-band design on 5 mil alumina (ε r  =9.9), and a W-band design on 5 mil z-cut quartz (ε r  =4.7) are discussed below. Models of these two designs were simulated using 3D FEM simulators, employing a relatively strict convergence criteria. S-parameter measurements of the transitions were facilitated by employing two identical transitions fixed in a back-to-back arrangement (as shown for example in FIG. 3, where the two transitions would be connected through a 50 ohm microstrip line, rather than the active MMIC devices shown). The transitions are connected using a 50 Ohm microstrip line, 955 mils long for the Q-band fixture and 830 mils long for the W-band fixture, to allow the distinct characterization of the transitions without any interactive effects. 
     FIG. 6 shows the theoretical values of: 
     |S 11  | dB  (Reference 90), |S 22  | dB  (Reference 92) and 
     |S 21  | dB  (Reference 94) 
     for the Q-band transition. Indicator 108 indicates that curves 110 and 112 use the leftmost ordinate values. Reference 90 which is curve 110 represents the reflection coefficient from the waveguide; reference 92 which is curve 112 represents the reflection coefficient from the microstrip line; and reference 94 which is curve 116 represents the transmission characteristics. Indicator 114 indicates that curve 116 uses the rightmost ordinate values. Theoretical dielectric and planar conductor losses are accounted for in the model simulation. The frequency rate is approximately in the 44 GHz region. For a 15 dB return loss, a bandwidth greater than 10% is predicted. The insertion loss of the transition throughout the band of interest is ˜0.35 dB. 
     FIG. 7 shows the Q-band measured data of two back-to-back transitions obtained on an automated network analyzer (ANA). The measured results corresponding to one transition can be determined from the back-to-back transitions data. Curve 118 represents the insertion loss. Curve 120 represents reflection coefficient. By accounting for the microstrip line and test fixture losses based on separate measurements (1.8 dB/in and 0.2 dB, respectively, at 44 GHz), the return and insertion losses of one transition can be calculated. A 10% bandwidth is deduced for a 15 dB return loss, and the insertion loss per transition is found to be less than 0.3 dB. Around the center of the band, a return loss better than 22 dB has been obtained. 
     FIG. 8 shows the theoretical values for the W-band transition including loss. Curve 122 represents the insertion loss response. Curve 124 represents the output reflection coefficient. Curve 126 represents the input reflection coefficient. The frequency range is approximately in the 94 GHz region. For a 15 dB return loss bandwidth, an insertion loss better than 0.35 dB can be achieved. The W-band design was implemented on a lower permittivity substrate (z-cut quartz) for bandwidth considerations. The higher overall circuit Q in this frequency band leads to a narrower response than that at Q-band. 
     FIG. 9 shows the W-band back-to-back transitions measured data. Curve 128 represents insertion loss. Curve 130 represents input reflection coefficient. From these, the frequency response of the transitions exhibits a relatively wider and flatter bandwidth than that shown in FIG. 8. A 12% bandwidth with a 15 dB return loss can be deduced. The insertion loss is found to be less than 0.2 dB per transition, using a value of 1.61 dB/in for the microstrip line and test fixture losses at 94 GHz. 
     The embodiments which have been set forth above were for the purpose of illustration and were not intended to limit the invention. It will be appreciated by those skilled in the art that various changes and modifications may be made to the embodiments discussed in the specification without departing from the spirit and scope of the invention as defined by the appended claims.