Abstract:
A feedback loop corrects timing errors by reducing deviations from a constant radar sweep rate. Errors are detected and fed back to a phase corrector in a high gain feedback system. A precision radar rangefinder can be implemented with a direct digital synthesizer (DDS) that includes feedback error correction for reducing range errors by, for example, 100 times, or to 0.1 mm. An error-corrected DDS swept timing system can enable a new generation of highly flexible, repeatable and accurate radar, laser and guided wave rangefinders.

Description:
BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The present invention relates to radar timing circuits and more particularly to precision swept delay circuits for expanded time ranging systems. It can be used to correct errors in a swept-delay clock for sampling radar, Time Domain Reflectometry (TDR) and laser systems. 
   2. Description of Related Art 
   High accuracy pulse-echo ranging systems, such as wideband and ultra-wideband pulsed radar, pulsed laser rangefinders, and time domain reflectometers, sweep a timing circuit across a range of delays. The timing circuit controls a receiver sampling gate such that when an echo signal coincides with the temporal location of the sampling gate, a sampled echo signal is obtained. The echo range is then determined from the timing circuit, so high accuracy timing is essential. A stroboscopic time expansion technique is employed, whereby the receiver sampling rate is set to a slightly lower rate than the transmit pulse rate to create a stroboscopic time expansion effect that expands the apparent output time by a large factor, such as 100,000. Expanded time allows vastly more accurate signal processing than possible with realtime systems. 
   A common approach to generate accurate swept timing employs two oscillators with frequencies F T  and F R  that are offset by a small amount F T −F R =Δ. In a ranging application, a transmit clock at frequency F T  triggers transmit pulses, and a receive clock at frequency F R  gates the echo pulses. If the receive clock is lower in frequency than the transmit clock by a small amount Δ, the phase of the receive clock will slip smoothly and linearly relative to the transmit clock such that one full cycle is slipped every 1/Δ seconds. Such a clock system forms a swept phase clock system. The term phase can also relate to time, since phase is another way to express time difference between the two clocks. Typical parameters are: transmit clock F T =2 MHz, receive clock F R =1.99999 MHz, frequency offset Δ=10 Hz, phase slip period=1/Δ=100 milliseconds, and a time expansion factor of F T /Δ=200,000. This two-oscillator technique was used in the 1960&#39;s in precision time-interval counters with sub-nanosecond resolution, and it appeared in a short-range radar in U.S. Pat. No. 4,132,991, “Method and Apparatus Utilizing Time-Expanded Pulse Sequences for Distance Measurement in a Radar,” by Wocher et al. 
   There are many influences that can affect the accuracy of the phase slip, including: (1) oscillator noise due to thermal and flicker effects, (2) transmit-to-receive clock cross-talk, and (3) thermal transients that typically do not track out between the two oscillators. The receive oscillator is typically locked to the offset frequency by a phase locked loop (PLL) circuit, which does a reasonable job when the offset frequency is above several hundred Hertz. Unfortunately, precision long range systems require extremely high accuracy, on the order of picoseconds, at offset frequencies on the order of 10 Hz. A PLL system cannot meet this requirement for the simple reason that the PLL loop response must be slower than 1/Δ, or typically slower than 100 ms, which is far too slow to control short term phase errors between the two clocks. 
   U.S. Pat. No. 6,404,288 to Bletz et al addresses the problems associated with controlling low offset frequencies by introducing three additional oscillators into a system further comprised of seven counters and two phase comparators, all to permit PLL control at higher offset frequencies than the final output offset frequency, which is obtained by frequency down-mixing. This system is too complex for many commercial applications and like the prior art, it does not control instantaneous voltage controlled oscillator (VCO) phase errors and crosstalk. 
   Swept timing can also be implemented using analog sweep techniques. Analog approaches to swept timing include: (1) an analog voltage ramp that drives a comparator, with the comparator reference voltage controlling the delay, or (2) a delay locked loop (DLL), wherein the delay, or phase, between transmit and receive clocks is measured and controlled with a feedback loop. Examples of DLL architectures are disclosed in U.S. Pat. No. 5,563,605, “Precision Digital Pulse Phase Generator” by the present inventor, Thomas Edward McEwan, and in U.S. Pat. No. 6,055,287 “Phase-Comparator-Less Delay Locked Loop”, also by the present inventor. The analog approaches are subject to component and temperature variations, and often require calibration during manufacture. There can also be accuracy limitations. 
   A radar timing system employing a direct digital synthesizer (DDS) is disclosed in U.S. patent application Ser. No. 11/351,924, “Direct Digital Synthesis Radar Timing System” by the present inventor. A DDS generates frequencies by digitally accumulating phase in a manner that directly emulates the definition of frequency. Frequency ω can be defined by a rate of change in phase φ or ω=φ/t, where t is time. Direct digital synthesis emulates this process by continually incrementing a digital phase value in discrete phase increments in a phase accumulator. It performs the accumulation in discrete time steps. The size of the discrete phase increment is set by a digital tuning word, and the discrete time steps are set by a DDS clock. Together, both define the synthesized frequency. This technique works well for low synthesized frequencies relative to the DDS clock frequency since a large number of small phase increments can be added in the phase accumulator to produce one full cycle spanning 0 to 2 π in phase, and a very smooth progression in phase can be realized. It does not work as well at higher frequencies that are required for radar. 
   In a radar system, a DDS drives a receive sampling gate at a frequency that is offset from a transmit pulse frequency to produce an expanded time sampled echo signal. The frequency offset generates a smoothly slipping phase between realtime received echoes and the sampling gate that stroboscopically expands the apparent time of the sampled echoes with an exemplary factor of 1-million and a range accuracy of 1-centimeter. The flexibility and repeatability of the digitally synthesized timing system is a quantum leap over analog prior art. However, the accuracy of currently available DDS chips is limited to about 0.05% of full scale range. Many applications require higher accuracy but cannot take advantage of the benefits of a DDS timing system due to limited accuracy. 
   A rate locked loop (RLL) timing system is disclosed in U.S. patent application Ser. No. 11/343,049, “Rate Locked Loop Radar Timing System” by the present inventor. An RLL regulates phase slip between two clock signals to provide precision timing for radar, TDR and laser ranging systems. A phase detector converts clock phase to voltage and the voltage is differentiated to provide a rate-of-change signal to a loop controller that precisely regulates the rate-of-phase change. The RLL controls a VCO to produce a constant, linear phase slip having phase errors below the time equivalent of 1-picosecond. However, the RLL lacks the repeatability and programmability of a digital timing system such as a DDS. 
   SUMMARY OF THE INVENTION 
   The present invention overcomes the limitations of the various analog and digital timing techniques used to generate a swept phase clock by employing an error correction feedback loop that reduces deviations from a constant sweep rate. Errors, i.e., deviations are detected and fed back to a phase corrector in a high gain feedback system. 
   A radar timing system having a constant sweep rate can be implemented with a direct digital synthesizer (DDS) that generates a receive clock that is offset from a transmit clock However, DDS timing errors can introduce 1 cm range error. A feedback error correction loop can reduce errors to less than 0.1 mm. The flexibility, repeatability and accuracy of an error-corrected DDS timing system can enable a new generation of highly accurate radar, laser and guided wave rangefinders. 
   The present invention provides an error-corrected swept phase timing system for expanded time radar that can include:
         a clock generator for providing a first clock signal and a swept phase signal,   a phase corrector including a control port for producing a second clock signal in response to the swept phase signal,   a phase detector for producing a phase detector output proportional to the phase between the first and second clock signals,   an error detector for producing an error signal proportional to phase errors between the transmit clock signal and the swept phase signal; and   a controller coupled to the control port for producing a control signal proportional to the error signal, wherein the first and second clock signals provide error corrected radar timing.       

   The error detector can comprise a first differentiator for producing a first derivative signal from the phase signal and a second differentiator for producing an error signal from the first derivative signal. It can also comprise a reference ramp generator for producing a reference ramp signal and a differencing element for producing an error signal proportional to the difference between the reference ramp signal and the phase signal. 
   The clock generator can include a reference oscillator to provide the first clock signal and a voltage controlled oscillator (VCO) to provide the swept phase signal. It can also include a reference oscillator to provide the first clock signal and a phase sweep circuit responsive to a range ramp to provide the swept phase signal. It can further include reference oscillator and a digital counter to provide the first clock signal and a direct digital synthesizer (DDS) provide the swept phase signal. The phase detector can consist of a flip-flop and a lowpass filter, the second differentiator can consist of an AC coupling circuit and the second clock signal can be harmonically related to the first clock signal. 
   The present invention can be used in expanded time radar, laser, and TDR ranging systems having high stability, flexible programmability, excellent repeatability and manufacturability, and an uncorrected phase accuracy on the order of 0.004 degrees using, for example, currently available, low cost DDS chips. Applications include pulse echo rangefinders for tank level measurement, environmental monitoring, industrial and robotic controls, digital handwriting capture, imaging radars, vehicle backup and collision warning radars, and universal object/obstacle detection and ranging. 
   A beneficial embodiment of the present invention is to provide a precision radar timing system that generates a highly accurate and repeatable phase slip to produce accurate radar signal time expansions and corresponding ranging accuracies. A further beneficial embodiment is to provide precision radar timing that is digitally and rapidly programmable. An even further beneficial embodiment of the present invention is to provide precision radar timing system that is highly reproducible, inherently calibrated and highly accurate. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a diagram of an error-corrected swept phase timing system of the present invention. 
       FIG. 2   a  is a diagram of a two oscillator dock generator. 
       FIG. 2   b  is a diagram of a single oscillator clock generator including a phase sweeper. 
       FIG. 2   c  is a diagram of a DDS based clock generator. 
       FIG. 3   a  is a diagram of a phase detector. 
       FIG. 3   b  is a diagram of a phase detector for harmonically related clocks. 
       FIG. 4  is a diagram of a phase corrector. 
       FIG. 5  is a diagram of a derivative circuit and a controller. 
       FIG. 6   a  is an error plot for a timing system without error correction (PRIOR ART). 
       FIG. 6   b  is an error plot for a timing system with error correction. 
       FIG. 7  is a diagram of the present invention in a ranging system. 
       FIG. 8  is a diagram of an error-corrected swept phase timing system including an alternative error detector. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   A detailed description of the present invention is provided below with reference to the figures. While illustrative component values and circuit parameters are given, other embodiments can be constructed with other component values and circuit parameters. All U.S. patents and copending U.S. applications cited herein are herein incorporated by reference. 
   General Description 
   The present invention overcomes the accuracy limitations of a DDS based clock generator and other clock generators by correcting errors in phase slip on a continuous and instantaneous basis. A beneficial example embodiment, as disclosed herein, employs a phase detector coupled directly between radar transmit and receive clocks, rather than through counter chains that are customary in PLL circuits, to produce a voltage proportional to instantaneous phase. When the phase between the clocks slips at a constant rate, because of the frequency offset between them, the phase detector output is a linear voltage ramp that increases with increasing phase values between 0 and 2 π and then it resets to 0 at 2 π, i.e., at the phase wrap point. The voltage ramp repeats at the offset frequency Δ. The voltage ramp is differentiated by a derivative circuit to produce a constant voltage proportional to the slope of the ramp, which can be termed the derivative voltage. The derivative voltage is applied to another derivative circuit that strips away the constant voltage produced by the first derivative circuit and allows only deviations in voltage from the first derivative circuit to pass as an error signal. A feedback controller controls a phase corrector in response to the error signal with the effect that phase errors are reduced, i.e., corrected. The amount of correction is a function of loop gain. If the slip rate varies, i.e., deviates, the high gain feedback controller instantaneously corrects the deviations. 
   The second derivative circuit outputs deviations in the rate of phase change. For a perfectly linear phase sweep, the second derivative circuit produces a zero error signal. The rate of sweep sets the output level from the first derivative circuit. The second derivative circuit rejects this output level so the system is not directly influenced by the sweep rate itself. The system is responsive to deviations in rate of change in phase and not to phase itself or to rate of change in phase. Consequently, the overall loop functions as an error corrector. High accuracy swept timing can be realized with low accuracy sweep systems when they are combined with the error corrector. Error correctors can be cascaded for increased error reduction. 
   One example swept timing system is based on a DDS as discussed in the Related Art section. The accuracy of a DDS timing system is limited by residual phase errors related to the number of accumulator bits in the DDS, which can be fairly large, e.g., 34 bits, and by sine ROM and DAC bit width, which can be 10-14 bits. The least significant bits (LSBs) from the accumulator are truncated to match the bit width of the DAC. A DDS in combination with a sine ROM, a DAC, and a reconstruction filter can provide an offset clock frequency having sufficiently small phase increments for sampling type stroboscopic radars having a full scale range error of 0.05%. When combined with the error corrector of the present invention, accuracy can be reduced to less than 0.001%. 
   Specific Description 
   Turning now to the drawings,  FIG. 1  illustrates an exemplary configuration of an error-corrected swept phase timing system  10  for expanded time radar of the present invention. A clock generator  12  provides a clock signal (CLK 1 ) on line  14  and a swept phase signal on line  16 . The swept phase signal is coupled to a phase corrector  32  that outputs a second clock signal (CLK 2 ) on line  20 . CLK 1  can be a transmit clock and CLK 2  can be a receive clock. The receive dock in a radar can be swept, but the transmit clock can be swept instead. 
   A phase detector  22  compares the phase between CLK 1  and CLK 2  and outputs a voltage V(φ) that is proportional to the CLK 1 -CLK 2  phase. Voltage V(φ) can have a ramp waveform, termed a phase ramp, when the CLK 1 -CLK 2  phase changes at a constant rate. 
   A first differentiator  24  differentiates V(φ)to produce a derivative voltage V′(φ) proportional to the rate-of-change in phase between CLK 1  and CLK 2 . Voltage V′(φ) is constant when V(φ) changes at a linear rate, representing a constant phase slip. A second differentiator  26  differentiates voltage V′(φ) to produce a second derivative voltage V″(φ). Voltage V″(φ) is an error signal representing deviations from a constant phase sweep rate. Controller  28  amplifies V″(φ) and produces a control voltage Vc proportional to V″(φ). Voltage Vc is applied as a negative feedback signal to a phase control port of phase corrector  32 , which controls the phase of CLK 2  relative to its input on line  16 . 
   Blocks  22 ,  24 ,  26 , and  28 , as shown in  FIG. 1 , form a high gain, high bandwidth continuous-mode feedback loop. Since the loop contains phase detector  22  and derivative elements  24  and  26 , it controls a second derivative of phase, or deviations in rate-of-change in phase. Accordingly, the feedback loop controls, i.e., corrects, phase deviations from a constant sweep. 
     FIG. 2   a  depicts an exemplary clock generator  12  having an independent reference oscillator  40 , which can be a quartz crystal oscillator that may be temperature compensated (TCXO) or ovenized for greater stability. Oscillator  40  operates at a frequency of Fref. VCO  42  produces a frequency that is offset frequency from Fref. The frequency offset causes the phase of oscillator  42  to slip relative to the phase of oscillator  40 , thereby producing a swept phase signal. A frequency control input adjusts the VCO frequency using a PLL or other control system. VCO  42  can be a quartz crystal oscillator with a varactor phase/frequency control element. 
     FIG. 2   b  depicts another exemplary dock generator  12  based on a single oscillator  40 , which directly provides CLK 1 . The swept phase signal is provided by a phase sweeper  44 , which is coupled to the CLK 1  line. The phase sweeper sweeps its output phase on line  16  in response to a ramp voltage. The maximum phase sweep range is normally limited to less than ½ π, which is sufficient for radar. 
     FIG. 2   c  depicts another exemplary clock generator  12  based on a single oscillator  40 , which provides CLK 1  on line  14  after division by N using counter  46 , where N can be an integer or an integer ratio. A DDS  48  is docked by oscillator  40 . The DDS produces an output frequency that is set by a digital tuning word. The tuning word can be set to cause the DDS to output a frequency that is offset from a sub-multiple of Fref. Filter  49  removes spurious frequency components from the DDS output and provides the swept phase signal on line  16 . Exemplary parameters for the CLK 1  frequency can be 2.000000 MHz and the swept phase signal frequency can be 1.999990 MHz. The difference frequency is 10 Hz and the swept phase signal slips at a smooth rate repeating at a 10 Hz rate. Once every 1/10 second, the phase of the CLK 1  and the swept phase signals align so there is zero phase between them for an instant. 
     FIG. 3   a  is an exemplary phase detector  22 , as shown in  FIG. 1 , that is based on a D-input latch (or flip-flop)  50 . Latch  50  is cleared by CLK 1  via edge coupling network  52 . After clearing, the next CLK 2  edge sets latch  50  so that the duty cycle of the Q output is proportional to the phase between CLK 1  and CLK 2 . Low pass filter  54  averages the duty cycle into a voltage V(φ) proportional to phase. 
     FIG. 3   b  depicts a further example of a phase detector wherein the CLK 1  signal is frequency divided by an integer N in counter  56 , such that V(φ) is proportional to the phase between a sub-multiple of the CLK 1  frequency and the direct frequency of CLK 2 . Counter  56  output is CLK 1 ′ at a sub-multiple N of CLK 1 . When the CLK 1 ′ is at a logic 1, latch  50  remains cleared, and when CLK 1 ′ is at logic 0, the next trigger edge of CLK 2  sets Q high. Since CLK 2  occurs at a higher rate than CLK 1 ′, the Q output, which is also CLK 2 ′, ranges over less than 2 π. For N=4, the phase range is ¼ π, a desirable range for many ranging systems. Further details on this harmonic mode can be found in U.S. Pat. No. 6,072,427, “precision Radar Timebase Using Harmonically Related oscillators,” by Thomas E. McEwan, the applicant of the present invention. Two frequencies are harmonically related if one is a multiple of the other, or dose to a multiple of the other, i.e., offset by a small difference frequency from the harmonic frequency. 
     FIG. 4  is an exemplary phase corrector  32  that includes an RC network  66  coupled to a threshold element  68 , a logic gate. RC network  66  slows the swept phase signal risetime and voltage Vc on line  30  provides an offset voltage that is applied to the input of gate  68 . The exact time that gate  68  thresholds on its input is a function of its input offset voltage. Therefore the timing, i.e. the phase, of the swept phase CLK 2  signal on line  20  is controlled by Vc. 
     FIG. 5  is an exemplary implementation of differentiators  24 ,  26  and controller  28 , as shown in  FIG. 1 . Phase detector  22  output V(φ) is applied to differentiation capacitor  70 , also labeled d/dt, which is coupled to the input of a transimpedance amplifier that includes op amp  72  and feedback resistor  74 , forming, in combination with capacitor  70 , a classic differentiator. Diode  76  conducts during the phase wrap transition at the 2 π points, i.e., during the positive edges seen in V(φ) waveform  89 , and acts to speed settling to the next negative going ramp of V(φ). Op amp  72  outputs a substantially constant voltage V′(φ) proportional to the rate of change of V(φ). 
   Switch  78  is normally closed and couples V′(φ) to a second differentiation capacitor  80 , also labeled d/dt. Capacitor  80  differentiates V′(φ) and couples a derivative voltage V″(φ) to resistor  82  and op amp  84 . Capacitor  80  forms an AC coupled circuit. Voltage V″(φ) is an error signal representing deviations from a perfectly linear sweep. Op amp  84  is a control amplifier that greatly amplifies the error signal to provide a feedback control voltage Vc on line  30  to phase corrector  32 . Capacitors  75 ,  86  define the control loop bandwidth. Capacitor  80  need not necessarily form a perfect differentiator; it functions to block the DC voltage level of V′(φ). 
   Switch  78  is opened by a pulse applied to the dashed S control line of  FIG. 5  shortly before the phase wrap. Opening switch  78  blocks phase wrap glitches from coupling to the control op amp. Switch  78  closes shortly after the phase wrap. The S control pulse can be derived from V(φ). Phase wrap glitches can limit the timing accuracy. Switch  88  is normally open and can close in compliment to switch  78 . The closure of switch  88  resets the output of op amp  84  at the phase wrap point and then switch  88  opens to allow extremely large DC gain, which helps reduce phase errors. Exemplary op amps  72 ,  84  are Texas Instruments, Inc. TLV274 and switches  78 ,  88  are Motorola, Inc. CMOS analog switches 74HC4066. 
     FIG. 6   a  (PRIOR ART) plots phase error between CLK 1 ′ and CLK 2 ′ for an actual timing system using harmonically related clocks and the phase comparator of  FIG. 3   b . A DDS dock generator as depicted in  FIG. 2   c  is used. Errors are indicated as the temporal equivalent of 15 picoseconds per division across a sweep range of 154 ns. CLK 1 ′ is operated at 1.625 MHz and CLK 2  at 6.5 MHz in a harmonic system as described with reference to  FIG. 3   b . Hence the sweep range is 1/6.5 MHz=154 ns, which corresponds to a phase range of ¼ π. Maximum errors are on the order of +/−60 picoseconds, or about +/−0.04% of full scale range. 
     FIG. 6   b  is a plot of the phase error for the system of  FIG. 6   a  further including an exemplary error corrector as illustrated by the timing system  10  of  FIG. 1 . Range marker  90  can correspond to zero range and range marker  92  can correspond to the maximum range for a rangefinder implementation. Errors between markers  90 ,  92  are on the order of 1-picosecond, or less than 0.001% of full scale range. 
     FIG. 7  illustrates an exemplary pulse-echo rangefinder  100  incorporating timing error corrector  10 , as shown in  FIG. 1 , of the present invention. Clock generator  12  couples a transmit dock signal TX CLK to transmitter  110  and, via phase corrector  32 , a receive clock signal RX CLK to receiver  112 . TX CLK triggers transmit pulses and transmitter  110  radiates corresponding radio or optical transmit pulses. Alternatively, transmitter  110  transmits electrical pulses along a conductor in a time domain reflectometer. Receiver  112  receives echo pulses produced by the transmitter. RX CLK gates the receiver, causing it to sample echoes at the instant of gating. Samples are output from the receiver on line  114  in expanded time as the phase of RX CLK slips relative to TX CLK. The samples on line  114  may occur on a pulse-by-pulse basis, one for each pulse of RX CLK, or the samples may be integrated to form an integrated output representing many RX CLK cycles. Receiver  112  can include processing, in which case the output on line  114  represents a processed output arising from samples taken at timing instants defined by RX CLK. 
   Phase ramp voltage V(φ) can be optionally coupled to receiver  114  via line  116  to control a variable gain amplifier to compensate echo versus range loss. Other uses for phase ramp voltage V(φ) include detecting the phase wraps at 2 π for generating reset pulses, generating switch control pulses for controller  28 , or for providing an analog indication of range. Blocks  22 ,  24 ,  26 ,  28  and  32  form a timing error corrector, which provides precision timing for rangefinder system  100 . Transmitter  110  and receiver  112  may be fashioned to operate with a single radiator or lens, or in the case of TDR, may be coupled onto a single conductor, as known in the art. 
     FIG. 8  illustrates another exemplary approach to obtaining an error signal for error correction. Clock generator  12 , phase detector  22 , control  28 , phase corrector  32 , CLK 1  and CLK 2  are as described previously. Reference ramp generator  25  generates a voltage ramp that matches phase ramp V(φ). Reference ramp generator  25  can consist of an analog generator or a digital generator as can be provided by a counter and a digital-to-analog converter (DAC). The optional dashed line in  FIG. 8  connecting phase detector  22  to reference ramp generator  25  can provide synchronization between the phase ramp V(φ) and reference ramp voltage Vr, so they both reset simultaneously. This connection may also provide amplitude regulation so both V(φ) and Vr match in peak-to-peak amplitude. Differencing element  23  subtracts Vr from V(φ) and outputs a difference voltage, i.e., an error signal to control  28 . Control  28  amplifies the error signal to produce a control voltage Vc at the phase control port of phase corrector  32 , which corrects sweep phase errors. Control  28  can be AC coupled to strip off any DC offset in the error signal and to control only the deviations. 
   Phase detector  22 , ramp generator  25  and differencing element  23  form an error detector. Similarly, referring to  FIG. 1 , phase detector  22 , first differentiator  24  and second differentiator  26  form an error detector. 
   The use of the word “radar” herein refers to traditional electromagnetic radar that employs microwaves or millimeter waves, and it also refers to optical radar, i.e., laser rangefinders, as well as guided wave radar, wherein radar pulses are guided along a electromagnetic guide wire or other conductor, as in TDR. “Radar” includes monostatic and bistatic systems, as well as radars having a single antenna/transducer. The use of the phrase “offset frequency” generally refers to an offset frequency between 1 and 1000 Hz between transmit and receive clock signals. However, the scope of the invention also encompasses larger offsets as may be required in various applications. Changes and modifications in the specifically described embodiments, including changing to digital and software embodiments, can be carried out without departing from the scope of the invention which is intended to be limited only by the scope of the appended claims.