Abstract:
An echo canceller for bidirectional transmission on two-wire metallic subscriber lines in an integrated service digital network employing a filter positioned in the echo path having the property that a zero point is located so as to cancel the echo path transfer function attributable to the inductance component of the line coupling transformer.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to an echo canceller for bidirectional transmission on two-wire subscriber lines utilizing metallic subscriber lines in an integrated service digital network (ISDN). 
     2. Description of the Prior Art 
     In recent years, a number of field trials for the ISDN, which adopts a bidirectional transmission system with an echo canceller utilizing existing metallic subscriber lines for interfacing with subscribers, have been conducted in many countries. 
     Such an echo canceller circuit is usually coupled to subscriber lines through transformers to exchange transmission and reception signals. For a so-called echo signal, which is part of the transmission signal reflected into the receiver section of its own circuit, the usual method is to achieve a certain degree of attenuation with a hybrid circuit having a simple balancing circuit, and to completely remove further the echo signal at a suppression level of about 60 decibels (dB) or more with an echo canceller which generates and subtracts an echo replica with an adaptive filter having transmission symbols as its input. At this time, the required number of taps of the transversal filter is determined by the length of the inpulse response of the echo entered into the echo canceller section, so that an echo path equalizing filter to shorten this impulse response length is interposed somewhere between the transmission driver and the echo canceller section. Since the long tailing-off part of an impulse response (echo tail) contains many low frequency components, conventionally such a high-pass filter as will suppress the low frequency components is used as this echo path equalizing filter. Usually a fully D.C.-intercepting type high-pass filter is used to make the direct current (D.C.) loss infintely great. A similar arrangement is described by P. F. Adams et al in a paper entitled &#34;Long reach duplex transmission systems for ISDN access&#34; published in the Br Telecom Technol J. Vol. 2, No. 2, April issue, 1984, pp. 35-42. 
     However, when the recently proposed 2B1Q line codes represented by a random sequence with a D.C. spectrum are applied to an echo canceller provided with the above-mentioned high-pass type filter, the echo canceller generates an unremoved echo residual corresponding to the result of convolution of an echo impulse response component left afteer the (N+1)th tap on the time axis. For instance, if the tap number N is 30 or so and an inductance value is about 50 mH, the average power of this echo residual will be about -10 dB in the absence of echo path equalizing filter, or about -40 dB where an echo path equalizer filter of a full D.C. intercepting type is used. These values are much less than the usually required suppression level of -60 dB. Meanwhile, though it is conceivable to achieve an adequate level of suppression by increasing the tap number N, the tap number N of the echo path equalizing filter then would have to be 100 or more, which is unrealistic. 
     An object of the present invention is, therefore, to provide an echo canceller for eliminating a pole having a long time constant and thereby accelerating the attenuation of the echo tail by cancelling the pole of an echo path transfer function attributable to the inductance component of the line coupling transformer. 
     Another object of the invention is to provide an echo canceller capable of reducing the number of taps of a filter by cancelling the pole of said echo path transfer function. 
     In order to achieve the foregoing objects, in an echo canceller according to the present invention, there is arranged within the echo path a filter having such a zero point as will cancel the pole of the echo path transfer function attributable to the inductance component of the line coupling transformer. 
    
    
     The present invention will be described below in conjunction with the accompanying drawings, in which: 
     FIG. 1 is a circuit diagram illustrating an example of common hybrid circuit; 
     FIG. 2 is a diagram illustrating echo tail waveforms; 
     FIG. 3 is a block diagram illustrating the basic structure of the present invention; 
     FIG. 4 is a block diagram illustrating a first preferred embodiment of the invention; and 
     FIG. 5 is a block diagram illustrating a second preferred embodiment of the invention. 
    
    
     In the drawings, the same reference numerals denote the same or corresponding structural elements. 
     DETAILED DESCRIPTION OF THE INVENTION 
     Referring to a model structure of a hybrid circuit illustrated in FIG. 1, a transfer function H(S) from a line driver 12 to a reception point of a hybrid section is the sum of a component H 1  (S) sent by way of a line coupling transformer 11 side and another component H 2  (S) sent via a balancing network (Z B ) 13 side. Out of these components, the component H 2  (s) is quick in attenuation of impulse response where the balancing network 13 consists of a resistor or a usually conceivable three-element impedance, so that the component H 1  (S) predominantly deterines the waveform at or beyond time 10T (T is the baud cycle). The component H 1  (S) can be represented by the following equation. ##EQU1## where R S  is the transmit impedance on the two-wire side; Z o , the impedance as viewed from the circuit toward the line side; S, jω; and L, the inductance of the transformer. To evaluate equation (1) with respect to the low frequency range, in long line transmission where the amplitude of the echo tail cannot be ignored relative to the receive signal level, Z o  is greater than R S  irrespective of the gauge and length of the line, or the presence or absence of bridged tap, so that equation (1) can be simulated as follows. ##EQU2## Equation (2) represents the primary high-pass characteristic of τ=L/R S , so that it is seen that the pole of this equation determines the echo tail. The attenuation time constant of the echo tail is equal to τ=L/R S . For instance, if L=50 mH and R S  =135 Ω, τ=about 370 μs and, if T=12.5 μs, τ≈30T. This corresponds to the t&gt;30T portion of waveform (A) in FIG. 2 of the impulse response of the echo in the absence of an echo path equalizing filter. 
     Then, if this echo tail waveform is passed through a(1-e &#39;ST )filter (not shown), the resultant amplitude will be the difference between two waves of the original echo tail form 1T apart from each other, and will attenuate by (1-e -T/ τ) times. If the values of the foregoing example are applied, the attenuation will be by approximately 1/30. This corresponds to the part of t&gt;30T of waveform (B) shown in FIG. 2. Waveform (B), though smaller in amplitude than waveform (A), has the same time constant as that. 
     Referring now to FIG. 3, a basic structure of the invention includes a hybrid circuit 1, an echo canceller circuit 2 and an echo path filter 3. The hybrid circuit 1, connected to a two-wire subscriber line 15 by way of a line-coupled transformer 11, achieves conversion between two and four wires. Whereas various structures are available for a hybrid circuit, including that of an electronic circuit and that of a transformer, what is to be used herein has to be connected to the subscriber line 15 by the transformer 11. 
     The echo canceller circuit 2 is connected to the four-wire side of the hybrid circuit 1. For this echo canceller 2 can usually be used either a tranversal filter type or a memory type for bidirectional digital transmission on a two-wire subscriber line. 
     FIG. 3 shows an outline of the transversal type filter. For details of this filter, reference is made to U.S. Pat. No. 4,087,654, for instance. 
     The echo path filter 3 is connected between the hybrid circuit 1 and echo canceller circuit 2, has a transfer function R(S) and operates to accelerate the attenuation of the echo tail generated on account of the D.C. interception characteristic of the transformer 11 of the hybrid circuit 1. 
     FIG. 4 illustrates a first embodiment of the invention. Description of the hybrid circuit 1 and echo canceller circuit 3 is dispensed with here because they are already explained with reference to FIG. 3. In this embodiment, the echo path filter 2 is composed of a digital filter having a delay element 31, an adder 32 and a multiplier 33, and has a transfer function of R(S)=1-(1-R S  /LT)e -ST . This echo path filter 3 and the hybrid circuit 1 are connected to each other through an A/D (analog to digital) converter 4. 
     The echo path filter 3 receives the output of the A/D converter 4. The output of the converter 4 is inputted to the delay circuit 31 and adder 32. The multiplier 33 multiplies an output from the delay element 31 and 1-R S  /LT to produce a transfer function (1-R S  /LT)e -ST . The adder 32 adds the output of the multiplier 33 and that of the A/C converter 4, and outputs the transfer function R.sub.(s) =1-(1-R S  /LT)e -ST  to the echo canceller circuit 2. 
     Incidentally, the echo path filter 3 may be composed of a memory and a microprocessor instead of the delay element 31, adder 32 and multiplier 33, and be controlled with a microprogram. In this case, the microprocessor may also be used for arithmetic operation of the echo canceller circuit 2. 
     The transfer function R(S) of the echo path filter 3 is in the form of 
     
         R(S)=1-(1-R.sub.S /LT)e.sup.-                              (3) 
    
     which can be approximated as follows if e -ST  is developed in the low frequency range: 
     
         R(S)≈R.sub.S /L+ST                                 (4) 
    
     
         provided that L&gt;&gt;R.sub.S T 
    
     From approximate equation (4), it is known that R(S) has the zero point at S=-R S  /L, and its frequency is the same as the pole of equation (2) above. Waveform (C) of FIG. 2 is the outcome of waveform (A) having passed this echo path filter 3. The transfer function R(S) has the advantage of eliminating waveforms having long time constants because of the cancellation of the pole of equation (2) at the zero point. As a result, waveform (C) attenuates quickly, becoming smaller in amplitude than waveform (B) at and after time 20T. 
     With respect to waveform (C), the number of taps of the echo canceller circit being supposed to be N, calculation of the echo residual resulting from the convolution of the impulse response component after (N+1)T with the transmission symbol sequence reveals that a suppression level of 60 dB or more can be achieved even at N=30, so that an echo canceller with a sufficient suppression level can be realized with a smaller number of taps than with waveform (A) or (B). 
     Incidentally, as the inductance of the transformer is usually so selected as to make L/R S  greater than T, 1-R S  /LT is a constant close to but smaller than 1. The pole attributable to the transformer inductance, as in equation (2) above, is S=-R S  /L in a long line condition, and where the transmission distance is different, some constant other than K=1-R S  /LT may be the optimum. 
     Referring to FIG. 5, a second embodiment of the invention includes the echo path filter 3 realized with an analog filter comprising resistors R1 and R2 and a capacitor C. In this case, the transfer coefficient R(S) is altered by the resistor R1 and R2 and the capacitor C as represented by equation (5). ##EQU3## When the pole of equation (2) and the zero point of equation (5) coincide with each other, i.e., when 
     
         R.sub.S /L=1/CR.sub.1 =ω.sub.0                       (6) 
    
     the overall characteristic of the echo path will have no component of a long time constant, and the impulse response of echoes will manifest a quickly attenuating characteristic, substantially similar to that of FIG. 2(C). Like in the example of FIG. 4, ω 0  =R S  /L will prove the optimum under a long line condition.