Abstract:
A highly linear, low-distortion H-bridge amplifier is described. The amplifier includes four interconnected power transistors connected to an inductive load. Each transistor has a precision voltage clamp connected between the source and drain to suppress ringing oscillations. Two or more of the transistors also include high-speed transient-voltage suppressers connected in parallel with their respective voltage clamps. These transient-voltage suppressers do not have the clamping accuracy of the voltage clamps, but respond much more quickly to suppress noise spikes. The amplifier therefore takes advantage of both the fast response time of the transient-voltage suppressers and the precise voltage-clamping characteristics of the voltage clamps.

Description:
BACKGROUND 
     1. Field of the Invention 
     This invention relates to control systems and methods for controlling inductive loads. More particularly, the invention relates to a low-noise and highly linear output stage for pulse-width-modulation (PWM) amplifiers. 
     2. Description of Related Art 
     Switching servo amplifiers are commonly used to supply drive current to inductive loads, such as linear, voice-coil, and DC motors. Such amplifiers often employ a type of output stage commonly known as H-bridge amplifiers, or simply “H-bridges. ” 
     FIG. 1 depicts a conventional H-bridge  100 . H-bridge  100  includes four transistors M 1 -M 4  configured to drive an inductive load  110 . Each transistor M 1 -M 4  has a corresponding diode D 1 -D 4  connected, in a reverse-current direction, from source to drain. Diodes D 1 -D 2  are typically fabricated integrally with respective transistors M 1 -M 4 . The gate voltages of transistors M 1 -M 4  are controlled by driver-amplifier circuits (not shown) of a conventional switching servo amplifier to alternate the direction of current flow through load  110 . Turning transistors M 1  and M 4  on and M 2  and M 3  off causes current to flow in one direction; turning transistors M 1  and M 4  off and M 2  and M 3  on causes current to flow in the other direction. Alternating between transistor pairs causes each terminal of load  110  to alternate between ground potential and the voltage level on power terminal +HV. The transistors are not switched simultaneously: some small delay ensures that transistors M 1  and M 2  (and similarly M 3  and M 4 ) do not conduct at the same time. 
     Conventional H-bridges and their associated circuitry are well known. An explanation of their operation is therefore omitted for brevity. For further information explaining the operation of several conventional H-bridge configurations, see the following U.S. Patents, the contents of which are incorporated herein by this reference: 
     4,581,565 to Van Pelt, et al., issued Apr. 8, 1986; 
     4,851,753 to Hamilton, issued Jul. 25, 1989; 
     4,873,618 to Fredrick et al., issued Oct. 10, 1989; 
     5,552,683 to Dargent, issued Sep. 3, 1996; and 
     5,596,446 to Plesko, issued Jan. 21, 1997. 
     Conventional H-bridge circuits are too noisy and produce excessive distortion for some precision applications. One facet of this noise is a “ringing” of the voltage levels on load terminals for a time after one pair of transistors is switched on and the alternate pair is switched off. 
     FIG. 2 illustrates the ringing of an output voltage level on terminal  120  of conventional H-bridge  100 . The first segment of the waveform (e.g., T 0  to T 1 ) represents the time during which transistors M 3  and M 2  are on, causing the voltage on terminal  120  to approach ground potential (e.g., zero volts). 
     Transistors M 3  and M 2  are switched off and transistors M 1  and M 4  switched on at time T 1 . Upon switching, the voltage on terminal  120  exceeds the supply voltage +HV for an instant due to the inductive “kick” of load  110  and associated leads. This noise spike is depicted as a spike  200 . Then, after some ringing  210 , the voltage on terminal  120  finally settles to the supply voltage +HV. H-bridge  100  also exhibits a noise spike  220  and ringing  230  at time T 2  when transistors M 3  and M 2  are switched on and transistors M 1  and M 4  are switched off. 
     Diodes D 1 -D 4  do much to limit the energy of spikes  200  and  220  and associated ringing  210  and  230 . Nevertheless, there remains a level of noise that is unacceptable for certain high-performance applications. There is therefore a need for a low-noise, highly linear output stage for pulse-width-modulation amplifiers. 
     SUMMARY 
     The present invention is directed to a highly linear PWM amplifier that exhibits low harmonic distortion. The amplifier includes four switching devices (e.g., power transistors) interconnected with each other and with an inductive load in an H configuration. Each transistor has a precision voltage clamp connected between the source and drain to suppress ringing oscillations. Two or more of the transistors also include ultra-high-speed transient-voltage suppressers connected in parallel with their respective voltage clamps. These transient-voltage suppressers do not have the clamping accuracy of the voltage clamps, but respond much more quickly to suppress noise spikes. An amplifier in accordance with the present invention therefore takes advantage of both the fast response time of the transient-voltage suppressers and the precise voltage-clamping characteristics of the voltage clamps. 
     This summary does not purport to define the scope of the invention. The scope of the invention is defined instead by the claims. 
    
    
     BRIEF DESCRIPTION OF THE FIGURES 
     The above and other features, aspects, and advantages of the present invention will become better understood with regard to the following description, appended claims, and accompanying figures, where: 
     FIG. 1 depicts a conventional H-bridge  100 ; 
     FIG. 2 shows a voltage signal exhibiting a ringing due to the inductive kick of a load; 
     FIG. 3 depicts an H-bridge circuit  300  in accordance with the present invention; 
     FIG. 4 depicts the voltage signal on one terminal of an inductive load  310  (solid line) superimposed on a relatively noisy prior-art voltage signal (dashed line); and 
     FIG. 5 is a detailed schematic diagram of H-bridge circuit  300 . 
    
    
     DETAILED DESCRIPTION 
     FIG. 3 depicts an H-bridge circuit  300  in accordance with an embodiment of the present invention. H-bridge  300  includes four MOSFET power transistors M 11 -M 14  interconnected with each other and to a pair of load terminals  302  and  304 . H-bridge  300  is configured to apply a high-voltage level and ground potential (e.g., 100 volts and zero volts) across an inductive load  310 . The applied voltages alternate between terminals  302  and  304  so that the current through load  310  periodically reverses direction. Transistors M 11 -M 14  are connected through conventional drivers  320  to a conventional switching servo amplifier (not shown). In one embodiment, driver circuits  320  are conventional MOSFET drivers available from Harris Semiconductor of Melbourne, Fla. as part number HIP2500IP. 
     Each of transistors M 11 -M 14  has a corresponding voltage-clamp diode VC 1 -VC 4  connected, in a reverse-current direction, from source to drain. Transistors M 12  and M 14  have respective transient-voltage suppressers TVS 2  and TVS 4  that, like voltage-clamp diodes VC 2  and VC 4 , are connected in a reverse-current direction from source to drain. 
     Voltage-clamp diodes VC 1 -VC 4  and transient-voltage suppressers TVS 2  and TVS 4  are selected to minimize the noise described above in connection with FIGS. 1 and 2. Transient-voltage suppressers TVS 2  and TVS 4  are selected for their fast response time. In one embodiment, TVS 2  and TVS 4  are available from General Instruments as part number 1.5KE100A. Those transient-voltage suppressers have a response time of approximately one picosecond, a minimum breakdown voltage of 95 volts, and a maximum breakdown voltage of 103 volts. 
     Voltage-clamp diodes VC 1 -VC 4  respond much more slowly than do transient-voltage suppressers TVS 2  and TVS 4 ; however, diodes VC 1 -VC 4  offer far greater voltage-clamping precision. In one embodiment, voltage-clamp diodes VC 1 -VC 4  are available from Motorola of Phoenix, Ariz. as part number MURS320T3. Those voltage-clamp diodes have a response time of approximately fifteen to thirty-five nanoseconds and a clamping voltage of one diode drop above or below a reference H-bridge circuits in accordance with the present invention take advantage of the best characteristics and transient-voltage suppressers TVS 2  and TVS 4  and voltage-clamp diodes VC 1 -VC 4  to minimize the noise associated with the inductive kick exhibited by load  310  during switching. Transistors M 11  and M 13  may have respective transient-voltage suppressers TVS 1  and TVS 3  connected in a reverse-current direction from source to drain to further reduce noise; however, some embodiments of the invention do not include transient-voltage suppressers TVS 1  and TVS 3 . 
     FIG. 4 depicts the voltage signal on terminal  302  of load  310  (solid line) superimposed on the prior-art voltage signal (dashed line) that includes spikes  200 / 230  and ringing  210 / 220  as discussed in connection with FIG.  2 . From time T 0  to T 1  transistors M 12  and M 13  are switched on and transistors M 11  and M 14  off. Terminal  302  is therefore pulled to ground and current flows from a high-voltage terminal +HV to ground via transistor M 13 , load  310 , and transistor M 12 . Next, at time T 1 , transistors M 12  and M 13  are switched off and transistors M 11  and M 14  switched on. The voltage on terminal  302  rises above the level of the high-voltage terminal +HV due to the inductive kick from load  310 ; however, transient-voltage suppresser TVS 2  conducts at a voltage level only slightly above the level of high-voltage terminal +HV, and thereby reduces or eliminates noise spike  200 . The fast response time of TVS 2  is necessary to suppress noise spike  200  because noise spike  200  occurs instantaneously. 
     From time T 1  to time T 2 , approximately fifteen to thirty-five nanoseconds, TVS 2  controls the voltage on node  330  to a level slightly above the voltage level on the high-voltage terminal +HV. The voltage on node  302  will vary from the voltage level on high-voltage terminal +HV by some error ξ due to the clamping-voltage error of TVS 2 . Voltage-clamp diode VC 1  is slower to respond but more accurate than is TVS 2 . Hence, at time T 2  voltage-clamp diode VC 1  clamps the voltage on node  302  to a level approximately one diode drop above the level on terminal +HV. The voltage on node  302  then settles to the level on terminal +HV after the excess energy from the inductive kick is fully absorbed. The voltage on terminal  302  then remains at approximately the level of terminal +HV until time T 3 , at which time transistors M 12  and M 13  are switched back on and transistors M 11  and M 14  switched off. 
     FIG. 5 is a detailed schematic diagram of an H-bridge  300  and associated drive circuitry configured in accordance with the present invention. H-bridge  300  is shown in FIG. 5 without transient-voltage suppressers TVS 1  and TVS 3  in accordance with one embodiment of the present invention. H-bridge  300  includes a pair of tank circuits  510  and  515  that absorb and dissipate energy from noise spikes. Similar circuits may be used for the ground side of H-bridge  300 ; however such circuits are typically not as necessary on the ground side because the ground terminal is designed to be of low impedance. Providing low impedance to ground is within the ordinary skill in the art. 
     Transistors M 11  through M 14  are selected to handle very high current. In an embodiment that requires transistors M 11  through M 14  to conduct two amps, for example, transistors M 11  through M 14  are rated to forward conduct 28 amps to avoid a large voltage drop across the transistors. This selection of transistors has been found to improve the noise characteristics of H-bridge  300  by reducing the source-to-drain voltage drop across transistors M 11  through M 14 . Suitable transistors are available from International Rectifier Corporation of El Segundo, Calif. as part number IRF540. A diode D 1  and capacitors C 1 -C 4  protect H-bridge  300  from potentially damaging voltage spikes and filter noise to and from the +HV power source. 
     While the present invention has been described in connection with specific embodiments, variations of these embodiments will be obvious to those of ordinary skill in the art. For example, conventional half-bridge amplifiers or multiple-phase amplifiers can be modified in accordance with the present invention for improved performance, or bipolar or other types of switches may be used. Moreover, some components are shown directly connected to one another while others are shown connected via intermediate components. In each instance the method of interconnection establishes some desired electrical communication between two or more circuit nodes. Such communication may often be accomplished using a number of circuit configurations, as will be understood by those of skill in the art. Therefore, the spirit and scope of the appended claims should not be limited to the foregoing description.