Abstract:
The disclosed current-controlled hysteretic oscillator operates by controlled currents opposing each other in differential pairs to set a controlled hysteresis for improved relaxation oscillations with immunity to phase or frequency error.

Description:
TECHNICAL FIELD 
     This subject matter is generally related to clocking and analog circuits. 
     BACKGROUND 
     Conventional oscillators and voltage controlled oscillators (VCOs) used in integrated circuit (IC) designs are susceptible to phase or frequency error and are limited in frequency range due to instability across a wide bandwidth. Commonly used ring oscillators are notoriously susceptible to phase or frequency error, altering the resonant frequency proportional to any phase error induced in the loop. Phase error can come from a number of sources, but a dominant source is noise on voltage supplies causing jitter and hence phase error on circuit elements in the loop. 
     Conventional approaches to deal with phase noise have been orientated toward controlling noise sources rather than designing oscillators to be more immune to such noise. One exception are oscillators that use higher-order filters to lock in frequency independent of phase error. But such conventional oscillators are expensive to implement, requiring more accurate discrete components, which is not amenable to IC design. Also, tuning to a tight band may not be useful in applications requiring wide frequency ranges, such as VCOs or oscillators requiring tuning across frequency. 
     SUMMARY 
     The disclosed current-controlled hysteretic oscillator operates by controlled currents opposing each other in differential pairs to set a controlled hysteresis for improved relaxation oscillations with immunity to phase or frequency error. The immunity to phase or frequency error allows tight locking to a desired frequency and an increase in stability across a wide range of frequencies. In some implementations, the disclosed oscillator provides about a 50% duty cycle across a wide range of frequencies. 
    
    
     
       DESCRIPTION OF DRAWINGS 
         FIG. 1  is a block diagram of an example current-controlled hysteretic oscillator. 
         FIG. 2  are plots of relative frequency range and stability for the circuit of  FIG. 1   
         FIG. 3  are plots of duty cycle across frequency and operating conditions for the circuit of  FIG. 1 . 
         FIG. 4  are plots of phase to frequency relative to ideal for the circuit of  FIG. 1 . 
     
    
    
     DETAILED DESCRIPTION 
       FIG. 1  is a block diagram of an example current-controlled hysteretic oscillator  100 . A control voltage V control  at the gate (or base if bipolar) of transistor M 2  sets the frequency of the oscillator  100 . The transistors M 1 , M 3 , form a first current mirror for mirroring current from M 2  to transistors M 5 , M 11  (e.g., PMOS transistors) in the upper portion of the oscillator  100  (hereinafter referred to as the “P” side), and the transistors M 3 , M 4  form a second current mirror for mirroring current from M 2  to transistors M 10 , M 16  (e.g., NMOS transistors) in the lower portion of the oscillator  100  (hereinafter referred to “N” side). 
     On the “P” side, a first branch of the oscillator  100  includes transistors M 5 , M 6 , M 7 , M 8 , M 9 , M 10 . Transistors M 6  and M 7  form a first differential pair. Current into the first differential pair is scaled by κβ contributed by transistor M 5 . The sizing of transistor M 5  is relative to β, set by the W/L ratio of M 5 , where W is channel width, L is channel length, and κ is a scalar less than 1 (e.g., 0.5 or 0.2) that defines the amount of hysteresis in the oscillator  100 . For example, a small value of κ provides less hysteresis than a large value of κ. 
     On the “N” side, transistors M 8  and M 9  form a second differential pair. Current into the second differential pair is scaled by κβ contributed by transistor M 10 . 
     On the “P” side, a second branch of the oscillator  100  includes transistors M 11 , M 12 , M 13 , M 14 , M 15 , M 16 . Transistors M 12  and M 13  form a third differential pair. Current into the third differential pair is scaled by β contributed by transistor M 11 . The sizing of transistor M 11  is relative to β, set by the W/L ratio of transistor M 1 . 
     On the “N” side, transistors M 14  and M 15  form a fourth differential pair. Current into the fourth differential pair is scaled by β contributed by transistor M 16 . Thus, the current into the third and fourth differential pairs is greater than the current into the first and second differential pairs due to the factor κ which is less than 1. 
     An operational amplifier A has a capacitor C across its positive and negative inputs. In some implementations, the capacitor C operates as a 1-pole filter. The size of V control , β, κ, and C determine the frequency operation, φ, where φ˜(1−κ)(V control −V T ) 2 A d β/(2V swing C), circuit constant A d  is the differential gain of amplifier A, V swing  is the magnitude of output signal swing, and V T  is the threshold voltage of the transistors. 
     The positive output of the amplifier A is coupled to node  2  and the negative output of amplifier A is coupled to node  4 . This effectively attaches a current sink (current to ground) to the non-inverting input to A (+A) and a current source (current from supply voltage) to inverting input to A (−A) when positive output is high and negative output is low. Conversely, a current source connects to the non-inverting input to A (+A) and a current sink to the inverting input to A (−A) when positive output is low and negative output is high. Both the current source and the current sink have equal magnitude and form the negative feedback currents to incite oscillation. 
     The positive output of the amplifier A is coupled to node  1  and the negative output of amplifier A is coupled to node  3 . This effectively attaches a current sink to the non-inverting input to A (+A) and a current source to inverting input to A (−A) when positive output is low and negative output is high. Conversely, a current source connects to the non-inverting input to A (+A) and a current sink to the inverting input to A (−A) when positive output is high and negative output is low. Both the current source and the current sink have equal magnitude and are a factor κ less than and opposite polarity to the negative feedback currents, thus forming hysteresis in conjunction with negative feedback currents. In this example configuration, the lesser hysteretic currents in the first branch will contend with the main feedback currents in the second branch oscillating in amplitudes relative to the amplifier A outputs. 
       FIG. 2  are plots of relative frequency range and stability for the circuit of  FIG. 1  and a conventional ring oscillator. The y-axis of the plots is frequency (MHz) and the x-axis of the plots is frequency voltage (Volts). In attempting to match frequency range, the greater instability inherent in ring oscillators is evident (e.g., a complete breakdown at fast conditions), and the duty cycle varies widely across frequency in contrast to the greater stability and constant 50% duty cycle inherent in the oscillator  100 . 
       FIG. 3  are plots of duty cycle across frequency and operating conditions for the circuit of  FIG. 1  and a conventional ring oscillator. The bold lines represent fast operating conditions and the thin lines represent slow operating conditions. Dashed lines represent nominal. In contrast to the oscillator  100 , the ring oscillator displays the characteristic unwieldiness with duty cycle across frequency. 
       FIG. 4  are plots of phase to frequency relative to ideal for the circuit of  FIG. 1 . A key advantage of the oscillator  100  is shown in  FIG. 4 . By better controlling phase via hysteresis (note linearity), and tightly controlled current, the nearer ideal performance of the oscillator  100  relative to the ring oscillator is discernable. The filters used on both oscillators are the most common and least expensive first-order low pass filters. By using n-order filters (where n is an integer greater than 1), the oscillator  100  can be improved, ideally approaching an impulse function as a band pass filter. However, given the first-order, low pass filters commonly used in IC oscillators, the oscillator  100  allows more ideal oscillator performance by controlling phase through hysteresis using the unique mirroring of currents to achieve the hysteresis, as described in reference to  FIG. 1 . 
     While this document contains many specific implementation details, these should not be construed as limitations on the scope what may be claimed, but rather as descriptions of features that may be specific to particular embodiments. Certain features that are described in this specification in the context of separate embodiments can also be implemented in combination in a single embodiment. Conversely, various features that are described in the context of a single embodiment can also be implemented in multiple embodiments separately or in any suitable subcombination. Moreover, although features may be described above as acting in certain combinations and even initially claimed as such, one or more features from a claimed combination can in some cases be excised from the combination, and the claimed combination may be directed to a subcombination or variation of a subcombination.