Abstract:
A combination of pre-distortion and post-distortion processes compensate for errors in I/Q channel orthogonality. The pre-distortion and post-distortion processes are calibrated to compensate for these errors at a variety of frequencies across a frequency span, thereby providing frequency-dependent compensation for I/Q channel mismatch. Pre-distortion calibration is effected by coupling the filtered analog I/Q modulated signals from the transmitter of a wireless transceiver directly to the analog-to-digital converters of the receiver of the wireless transceiver. Coupling the analog I/Q modulated signals from the transmitter directly to the channel filters that precede the analog-to-digital converters of the receiver effects post-distortion calibration.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     This invention relates to the field of communications, and in particular to a quadrature transceiver that includes pre-distortion and post-distortion compensation for frequency-dependent I/Q channel mismatch. 
     2. Description of Related Art 
     The use of quadrature modulation and demodulation is a common communication technique for communicating digital data as a stream of two-bit symbols. A first stream corresponding to one of the bits of the symbol is modulated by an “in-phase” (I) oscillation signal, and a second stream corresponding to the other bit of the symbol is modulated by a “quadrature-phase” (Q) oscillation signal that is ninety degrees out of phase from the in-phase (I) oscillation signal. The I and Q modulated signals are combined to form a composite signal for transmission. The orthogonal nature of the modulation allows for a reliable demodulation of individual I and Q modulated bit streams at a receiving system. 
     IEEE 802.11a/g specifies an Orthogonal Frequency Division Multiplex (OFDM) scheme that employs a combination of frequency division multiplexing and quadrature modulation and demodulation to effect high-speed wireless data transfer. At the OFDM transmitter, the outputs of a plurality of quadrature modulation systems are frequency-division-multiplexed for transmission to a corresponding OFDM receiver. As in all quadrature modulation and demodulation systems, OFDM systems are sensitive to phase shifts that cause the I and Q modulated signals to become non-orthogonal, commonly termed “I/Q channel mismatch”. 
     U.S. Pat. No. 6,298,035 “ESTIMATION OF TWO PROPAGATION CHANNELS IN OFDM”, issued Oct. 2, 2001 to Juha Heiskala, incorporated by reference herein, provides an overview of the principles of OFDM modulation and demodulation, and discloses a method of estimating the frequency response of each channel by transmitting select training symbols between two transceivers. This method particularly addresses the frequency-dependent effects caused by multipath fading and interference, and provides an adaptive solution based on actual transmissions from one transceiver to another. 
     BRIEF SUMMARY OF THE INVENTION 
     It is an object of this invention to provide a system and method that minimizes I/Q channel mismatch. It is a further object of this invention to provide a system and method for minimizing I/Q mismatch across a range of frequencies. It is a further object of this invention to provide a system and method that effects autonomous I/Q mismatch compensation within a single transceiver. 
     These objects, and others, are achieved by providing a combination of pre-distortion and post-distortion processes that compensate for errors in I/Q channel orthogonality. The pre-distortion and post-distortion processes are calibrated to compensate for these errors at a variety of frequencies across a frequency span, thereby providing frequency-dependent compensation for I/Q channel mismatch. Pre-distortion calibration is effected by coupling the filtered analog I/Q modulated signals from the transmitter of a wireless transceiver directly to the analog-to-digital converters of the receiver of the wireless transceiver. Post-distortion calibration is effected by coupling the analog I/Q modulated signals from the transmitter directly to the channel filters that precede the analog-to-digital converters of the receiver. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The invention is explained in further detail, and by way of example, with reference to the accompanying drawings wherein: 
     FIG. 1 illustrates an example block diagram of a transceiver with pre-distortion and post-distortion compensation for I/Q channel mismatch in accordance with this invention. 
    
    
     Throughout the drawings, the same reference numerals indicate similar or corresponding features or functions. 
     DETAILED DESCRIPTION OF THE INVENTION 
     This invention is premised on the observation that a substantial amount of I/Q channel mismatch can be introduced by the filters that are used at the transmitter to limit the bandwidth of the I/Q modulated output and the filters that are used at the receiver to isolated the transmitted I/Q modulated signal. This “locally-produced” I/Q mismatch is particularly acute in OFDM systems because of the filtering required at each of the plurality of quadrature modulation systems to avoid interference with each other, and because of the filtering required to isolate each of the frequency-division-multiplexed quadrature-modulated signals. 
     FIG. 1 illustrates an example block diagram of a transceiver with pre-distortion and post-distortion compensation for locally produced I/Q channel mismatch in accordance with this invention. The transceiver includes a conventional receiver section  100 , comprising the components whose reference numerals start with the digit “ 1 ”, a conventional transmitter section  200 , comprising the components whose reference numerals start with the digit “ 2 ”, and a calibration/ compensation section  300 , comprising the components whose reference numerals start with the digit “ 3 ”. 
     The receiver  100  includes a tunable front end  110  whose output is demodulated by a quadrature demodulator to provide quadrature output signals I and Q. For ease of illustration, only one branch of the quadrature demodulator is described herein, the other branch being functionally equivalent, but operating at an orthogonal phase provided by the quadrature phase generator  180 . The output of the front end  110  is demodulated by a mixer  120 , and filtered by a filter  130 . A tunable amplifier  140  provides a baseband analog signal, which is converted into digital samples via the analog-to-digital converter (ADC)  150 . 
     The transmitter  200  receives two digital streams for I and Q channel modulation and transmission. As with the receiver  100 , for ease of illustration, only one branch of the quadrature modulator is described herein, the other branch being functionally equivalent. A digital-to-analog (DAC) converter  220  converts the samples of the digital input stream into an analog signal that is filtered by the filter  230  and provided to a tuned amplifier  240 . The mixer  250  provides the quadrature modulation, the streams being mixed via modulation signals I and Q that are separated in phase by ninety degrees. The adder  260  combines the quadrature-modulated signals, and an amplifier  270  prepares the composite signal for transmission. 
     Of particular note are the filters  130  and  230 . These filters are bandpass filters that are configured to attenuate signals above a given cutoff frequency. As is known in the art, in addition to providing this frequency-dependent attenuation, filters generally introduce a frequency-dependent phase-shift. If the filters in each channel are identical, the phase-shift that is introduced will be of no consequence. In the receiver  100 , if the received signal includes orthogonal components, the quadrature mixers  120  will provide output streams that are in-phase with each other. If the filters  130  in each channel are identical, the phase-shift that is introduced to each stream will be identical, and the streams remain in-phase relative to each other. In the transmitter  200 , the input streams to each channel are assumed to be in-phase with each other. If the filters  230  in each channel are identical, the streams remain in-phase relative to each other, and the quadrature modulation via the mixers  250  provides output signals that are phase-shifted from each other by ninety degrees. 
     If either of the pairs of filters  130 ,  230  are not identical, the streams that are nominally in-phase with each other will exhibit a phase-shift relative to each other, particularly at or near the cutoff frequency of the filter, where the substantial frequency-dependent attenuation and frequency-dependent phase-shift are introduced in each channel. At the receiver  100 , if orthogonal signals are received, the input to each filter  130  will be in-phase with each other, but if the filters  130  in each channel are not identical, the output stream from the filters  130  will be out of phase with each other. If the phase-shift difference between the filters is substantial, the bit-stream output from the ADC  150  of the channels will be out of phase with each other. 
     As noted above, OFDM systems are particularly susceptible to frequency-dependent phase-shifts because of the sharp cutoffs required to minimize interference and to isolate transmitted signals. Although each filter pair  130 ,  230  is designed to comprise identical filters in each channel, the fabrication process can introduce unpredictable variations in the actual cutoff frequency realized by each filter. Because of the sharp filter response required, minor shifts in the cutoff frequency can introduce substantial phase-differences between the filters in each of the I/Q channels. 
     In accordance with this invention, an autonomous calibration/compensation system  300  is included within the receiver  100  and transmitter  200  of the transceiver. To compensate for the phase-shifts produced by the filter  230  in the transmitter  200 , a pre-distortion component  330  is provided that bit-phase-shifts one of the digital input streams such that the outputs from the filters  230  are in-phase with each other, corresponding to the assumed in-phase relationship between the two digital input streams. That is, as required, the pre-distortion component  330  delays one of the I or Q digital input streams, based on the analog-phase-shift that is produced by the filters  230 , as fabricated. Although an independent output processor could be used to determine the phase shift that is introduced by the transmit filters  230 , a preferred embodiment of this invention uses the output of the ADCs  150  to decode the analog outputs from the filters  230 . In-phase test signals are applied to the DACs  220  and filters  230 , and the degree of bit-shift that is introduced by the filters  230  and DAC  220  is determined by comparing the output of the ADCs  150  to the in-phase test signals. 
     To compensate for phase-shifts produced by the filter  130  in the receiver  100 , a post-distortion component  340  applies a bit-phase-shift to one of the I/Q digital output streams from the receiver  100  so that the streams are placed back in-phase, corresponding to their assumed in-phase condition from the mixers  120 . That is, as required, the post-distortion component  340  delays one of the I or Q output streams, based on the analog-phase-shift that is produced by the filters  130  as fabricated. Although an independent test signal could be generated to determine the required compensation, a preferred embodiment of this invention uses the “in-phase” channel signals from the transmitter  200 . Again, a test sequence is applied to the input of the transmitter  200 , and the output of the ADCs  150  are compared to the test sequence to determine the amount of bit-phase-shift that is caused by the analog-phase-shift introduced by the filters  130 . 
     A calibration controller  310  controls switches  320   a-b  to selectively engage the calibration process, and to selectively couple the analog signals from the transmitter to either the ADCs  150 , for calibration of the filters  230 , or to the filters  130 , for calibration of the filters  130 . In a preferred embodiment, the aforementioned test sequence that is provided to the transmitter  200  is configured to provide a calibration measure across a range of frequencies so as to allow for a frequency-dependent calibration and compensation method and system. 
     In a preferred embodiment, during transmitter calibration with the pre-distortion component  330  disabled, a signal I n =cos(ω n t) is applied to the I input, and Q n =sin(ω n t) is applied to the Q input for each of the sub-carriers of the OFDM signal, where ω n  is the n th  sub-carrier of the OFDM signal. As illustrated in FIG. 1, in the transmit-calibration mode, the output of the filters  230  from this applied input is fed directly to the ADC converters  150  in the receiver  100 . Defining R I  as the I output, and RQ as the Q output of the receiver  100 , the transmitter gain imbalance g T  and phase imbalance θ T  at each ω n  is given as:                g     T   n       =              R     Q   n         R     I   n                            and             (   1   )                 θ     T   n       =       arg        (       R     Q   n         R     I   n         )       .             (   2   )                                
     Designating I 0  and Q 0  as the input to the transmitter, I/Q imbalance at this ω n  can be represented in matrix form as:                [         I           Q         ]     =       A     T   n            [           I   0               Q   0           ]               (   3   )                                
     where                A     T   n       =       [         1           -     g     T   n            cos                   θ   n               0           g     T   n          cos                   θ               n        T               ]                   is                 the                 transmitter                 imbalance                   matrix   .               (   4   )                                
     To compensate for this transmitter I/Q imbalance, the inverse of the transmitter imbalance matrix A Tn  is applied at the pre-distortion component  330  at each ωn, wherein                A     T   n       -   1       =       [         1         tg        (     θ     T   n       )               0           1   /     g     T   n              cos        (     θ     T   n       )               ]     .             (   5   )                                
     In a preferred embodiment, during receiver calibration with the pre-distortion component  330  enabled to compensate for the transmitter I/Q imbalance, and the post-distortion component  340  disabled, a signal I n =cos(ω n t) is applied to the I input, and Q n =sin(ω n t) is applied to the Q input of the transmitter  200 , for each of the sub-carriers of the OFDM signal, where ω n  is the n th  sub-carrier of the OFDM signal. As illustrated in FIG. 1, in the receive-calibration mode, the output of the filters  230  from this applied input is fed directly to the filters  130  in the receiver  100 . Defining R I  as the I output, and R Q  as the Q output of the receiver  100 , the receiver gain imbalance g R  and phase imbalance θ R  at each ω n  is given as:                g     R   n       =              R     Q   n         R     I   n                            and             (   6   )                 θ     R   n       =       arg        (       R     Q   n         R     I   n         )       .             (   7   )                                
     Designating I 0  and Q 0  as the input to the transmitter, I/Q imbalance at this ω n  can be represented in matrix form as:                [         I           Q         ]     =       A     R   n            [           I   0               Q   0           ]               (   8   )                                
     where                A     R   n       =       [         1           -     g     R   n            cos                   θ     R   n                 0           g     R   n          cos                   θ     R   n               ]                   is                 the                 receiver                 imbalance                   matrix   .               (   9   )                                
     To compensate for this receiver I/Q imbalance, the inverse of the imbalance matrix A Rn  is applied at the post-distortion component  340  at each ωn, wherein          A     R   n       -   1       =       [         1         tg        (     θ     R   n       )               0           1   /     g     R   n              cos        (     θ     R   n       )               ]     .                            
     By applying the inverse of the transmitter I/Q imbalance at the pre-distortion component  330 , and the inverse of the receiver I/Q imbalance at the post-distortion component  340 , the effects of transmitter and receiver I/Q imbalance in a transceiver are minimized. 
     The foregoing merely illustrates the principles of the invention. It will thus be appreciated that those skilled in the art will be able to devise various arrangements which, although not explicitly described or shown herein, embody the principles of the invention and are thus within the spirit and scope of the following claims.