Abstract:
In an output stage of an operational amplifier, first and second transistors each provide a collector current under quiescent conditions to first and second current sources. A resistor receives a portion of one the collector currents and produces a resistor voltage in response. An output transistor provides a quiescent current having a value calculated as a function of the resistor voltage and a base-emitter voltage of the second transistor.

Description:
BACKGROUND 
   1. Field of the Invention 
   Aspects of the present invention relate generally to an operational amplifier, and more particularly to setting the quiescent current in a rail-to-rail output stage of an operational amplifier. 
   2. Description of Related Art 
   Operational amplifiers (“op-amps”) are widely known and used electronic devices that may be employed in a variety of applications. Generally, an op-amp is a DC-coupled high gain electronic voltage amplifier with differential inputs and a single output. As op-amps trend toward using lower supply voltages, the ability for the op-amp to swing from rail to rail has increased in importance. By being able to swing from rail to rail, the op-amp may use its full or close to its full voltage range on output and input. Rail-to-rail op-amps may be suited for low voltage applications. 
   However, random variations in the quiescent current flowing in the op-amp may occur due to manufacturing processes. These variations may affect the ability to properly bias an output transistor to control and maintain the desired quiescent current in a circuit. 
   Therefore, it may be desirable to provide a circuit arrangement that precisely sets the quiescent current in a rail-to-rail output stage of the op-amp. 
   SUMMARY 
   Embodiments of the present invention overcome the above-mentioned and various other shortcomings of conventional technology, providing an output stage of an op-amp having first and second transistors that each may provide a collector current under quiescent conditions to first and second current sources. A resistor may receive a portion of one of the collector currents and produce a resistor voltage in response. An output transistor may provide a quiescent current having a value calculated as a function of the resistor voltage and a base-emitter voltage of the second transistor. 
   The foregoing and other aspects of various embodiments of the present invention will be apparent through examination of the following detailed description thereof in conjunction with the accompanying drawing figures. 

   
     BRIEF DESCRIPTION OF THE DRAWING FIGURES 
       FIG. 1  is a schematic diagram of an embodiment of a portion of an output stage of an op-amp. 
       FIG. 2  is a schematic diagram of an embodiment of a portion of an output stage of an op-amp. 
       FIG. 3  is a schematic diagram of an embodiment of a portion of an output stage of an op-amp. 
       FIG. 4  is a schematic diagram of an embodiment of an output stage of an op-amp. 
   

   DETAILED DESCRIPTION 
   It will be appreciated from the following description that the embodiments set forth herein may have utility in connection with op-amps having various applications, including but not limited to industrial process control, battery-powered instrumentation, power supply control and protection, telecommunications, remote sensing, low voltage strain gage amplifiers, and DAC output amplifiers. 
   By way of illustration,  FIG. 1  is a schematic diagram of an embodiment of a portion of an output stage of an op-amp. Transistors Q 1   105  and Q 2   110  may be bipolar PNP transistors. Transistors Q 1   105  and Q 2   110  may be identical or matching transistors. The transistors Q 1   105  and Q 2   110  may be connected together at their respective bases. An emitter of Q 1   105  may be connected to a supply voltage rail, while a collector is connected to a current source I 1   130 . An emitter of transistor Q 2   110  may be connected in series to one end of resistor R 1   115  with the other end of resistor R 1   115  connected to the supply voltage rail. A collector of transistor Q 2   110  may be connected to a current source I 2   135 . Resistor R 1   115  may have its value set such that in quiescent conditions (i.e., with no input signal applied) resistor R 1   115  degenerates the base-emitter voltage V BE  of transistor Q 2   110  through the voltage drop across resistor  115 . In one embodiment, resistor R 1   115  may have a value of 200Ω. As a result of the voltage drop across resistor  115 , the base-emitter voltage V BE  of Q 2   110  may be smaller than the base-emitter voltage V BE  of Q 1   105 , and correspondingly, the collector current I CQ2  of transistor Q 2   110  may be smaller than the collector current I CQ1  of transistor Q 1   105 . Because collector current I CQ2  of Q 2   110  is smaller than collector current I CQ1  of Q 1   105 , a difference in collector currents, dI C , may exist between the two transistors Q 1   105  and Q 2   110  (i.e., I CQ1 −I CQ2 =dI C ). 
   In one embodiment, current sources I 1   130  and I 2   135  may demand equal amounts of current from transistors Q 1   105  and Q 2   110 . For example, current sources I 1   130  and I 2   135  may ask for 12 μA from transistors Q 1   105  and Q 2   110 . Because Q 2 &#39;s  110  collector current I CQ2  is less than Q 1 &#39;s  105  collector current I CQ1 , current source I 2   135  may obtain the difference in current dI C  from diode D 1   120 . In the embodiment where current sources I 1   130  and I 2   135  are demanding 12 μA from transistors Q 1   105  and Q 2   110 , 12 μA of current may be provided by the collector of transistor Q 1   105 , while only 11 μA of current may flow from the collector of transistor Q 2   110 . Diode D 1   120  may supply approximately 1 μA if base current errors from transistor Q 3   125  are ignored. Diode D 1   120  may mirror current dI C  to output stage transistor Q 3   125 , with transistor Q 3   125  having an 80:1 ratio relative to diode D 1 . Output stage transistor Q 3  may be a bipolar PNP transistor. The difference current dI C  mirrored by the diode to transistor Q 3   125  and amplified by transistor Q 3   125  sets the value of quiescent current I CQ3  in transistor Q 3   125 . Using the exemplary figures recited herein in the discussion of  FIG. 1 , the quiescent current may be approximately 80 μA if the current dI C  supplied by the diode D 1   120  is approximately 1 μA. 
   In the foregoing embodiment of  FIG. 1 , the quiescent current I CQ3  may be determined and set as a function of the value of degeneration resistor R 1   115 . To adjust the quiescent current I CQ3 , the value of the degeneration resistor R 1   115  may be changed. A drawback of using the degeneration resistor R 1   115  to set the quiescent current I CQ3  of transistor Q 3   125  is that the emitters of transistors Q 1   105  and Q 2   110  must perfectly match. For instance, if resistor R 1  has a value of 200Ω and current sources I 1   130  and I 2   135  both demand 12 uA, a 2.4 mV drop is formed on R 1 . Assuming no base current error and a perfect match between the emitters of Q 1   105  and Q 2   110 , Q 1 &#39;s collector current I C  would be about 10% larger than Q 2 &#39;s collector current I C . If process variation was bad enough to render Q 1 &#39;s  105  emitter 10% smaller than Q 2 &#39;s  110  emitter then, in effect, there would be no difference between Q 1 &#39;s collector current I C  and Q 2 &#39;s collector current I C  (i.e., dI C ≈0). 
     FIG. 2  is a schematic diagram of an embodiment of a portion of an output stage of an op-amp. As shown in  FIG. 2 , transistors Q 1   205  and Q 2   210  may be PNP bipolar transistors. However, in accordance with the principles of the invention, these transistors are not so limited. The emitters of both transistors Q 1   205  and Q 2   210  may be connected to the same rail. The bases of the respective transistors may be connected together. The areas of the emitters of respective transistors Q 1   205  and Q 2   210  may be mismatched, such that the emitter for one transistor has a larger area relative to the emitter for the other transistor. In one embodiment, the ratio of the areas of the emitters of Q 1   205  and Q 2   210  may be 5:6, but other ratios may be used as long as a difference in the base-emitter voltages of the two transistors exists. In one embodiment, this base-emitter voltage difference may be at least 1 mV. 
   A resistor R 1   215  may be connected at one end to the bases of the transistors Q 1   205  and Q 2   210  and the collector of Q 1   205 , and at the other end to the collector of transistor Q 2 . In one embodiment, resistor R 1   215  may have a value of 20 kΩ. The collector of transistor Q 2   210  and resistor R 1   215  may be connected to the base of transistor Q 3   230  and to current source I 2   225 . The collector of transistor Q 1   205  may be connected to current source I 1   220 . In one embodiment, current sources I 1   220  and I 2   225  may demand and supply equal amounts of current I. In one embodiment, the current I demanded by current sources I 1   220  and I 2   225  may be 12 μA. If the emitter of transistor Q 1   205  is ⅚th the size of the emitter of transistor Q 2   210 , collector current I CQ1  of Q 1   205  will be approximately ⅚th of collector current I CQ2  of Q 2   210 . The difference in current, dI C , between I CQ1  and I CQ2  (i.e., ⅙th of I) may be split between the two current sources I 1   220  and I 2   225  since the two current sources I 1   220  and I 2   225  are demanding equal amounts of current I from transistors Q 1   205  and Q 2   210 . Half of this current difference, dI C , that is 1 μA or 1/12 of the current I for the embodiment where I=12 μA, flowing from Q 2   210  to current source I 1   220  may flow through resistor R 1   215 , setting up a voltage drop dV BE  (dV BE =dI C *R 1 ) across resistor R 1   215 . The base-emitter voltage of transistor Q 3   230  may be expressed as V BEQ3 ≈V BEQ1 −dV BE , where dV BE = 1/12*I*R 1 . 
   A quiescent current I CQ3  of transistor Q 3   230  may have a value calculated as a function of the base-emitter voltage V BEQ3  of transistor Q 3   230 , with the value of the quiescent current in transistor Q 3   230  determined using the equations:
 
 V   BEQ3   =V   BEQ1 − 1/12 *I*R 1  (1)
 
 V   T  ln( I   CQ3 /80)= V   T  ln( I   CQ1 /5)− 1/12* I*R 1  (2)
 
 V   T  ln( I   CQ3 /(16* I   CQ1 ))=− 1/12 *I*R 1  (3)
 
 I   CQ3   =e   ((−1/12*I*R1)/26 mv) *16* I   CQ1   (4)
 
   In the equations above, the value 16 used in equation (3) may represent the ratio between transistor Q 3   230  and transistor Q 1   205 . V T  is the thermal voltage determined by the equation V T =kT/q, with k being Boltzmann&#39;s constant, T being the absolute temperature in Kelvins, and q being the magnitude of the electrical charge on the electron (in coulombs). V T  is approximately 26 mV. The base-emitter voltage V BEQ1  of transistor Q 1   205  may vary logarithmically with any variation in the current source I 1   220 . The voltage drop dV BE  across resistor R 1   215  may vary little because of the proximity and like diffusions of transistors Q 1   205  and Q 2   210 . 
   Compared to the embodiment of  FIG. 1  which relies on the absolute precision of the value of the degeneration resistor R 1   115  to set and control the quiescent current I CQ3  of transistor Q 3   125 , the circuit of  FIG. 2  may raise or lower the quiescent current I CQ3  of transistor Q 3   230  by changing the ratio of the emitter areas of transistors Q 1   205  and Q 2   210 . A larger difference in the ratio of emitter areas of transistors Q 1   205  and Q 2   210  may decrease the quiescent current flowing in transistor Q 3   230 , while a smaller difference in the ratio of the emitter areas of transistors Q 1   205  and Q 2   210  may increase the quiescent current. The embodiment of  FIG. 2  may further eliminate the need for the diode D 1   120  found in the embodiment of  FIG. 1  that mirrors the quiescent current to transistor Q 3 . The circuit of  FIG. 2  is inherently more stable than the embodiment of  FIG. 1 , with the risk of shutting off output stage transistor Q 3   235  greatly reduced, if not eliminated. 
   In an alternative embodiment, instead of mismatching the area of the emitters of transistors Q 1   205  and Q 2   210 , the circuit embodied in  FIG. 2  may have matching transistors Q 1   205  and Q 2   210 . The current sources I 1   220  and I 2   225  may demand different amounts of current from transistors Q 1   205  and Q 2   210 . For example, current source I 1   220  may demand 13 μA and current source I 2   225  may demand 11 μA. Because current source I 1   220  demands more current than current source I 2   225 , a portion of current supplied by transistor Q 2   210  may be redirected to feed current source I 1   220 , with the redirected portion of current routing through resistor R 1   215 , which in one embodiment may have a value of 20 kΩ. The resulting voltage drop, dV BE , across resistor R 1   215  may be used in conjunction with the base-emitter voltage V BE  of transistor Q 1   210  in calculating the base-emitter voltage V BEQ3  of transistor Q 3   230 . The base-emitter voltage V BEQ3  may be approximated as V BEQ1 −dV BE . The quiescent current I CQ3  of transistor Q 3   230  may be set and calculated as a function of the base-emitter voltage V BEQ3  of transistor Q 3   230  using the equations described above. 
     FIG. 3  is a schematic diagram of an embodiment of a portion of an output stage of an op-amp. In  FIG. 3 , transistors Q 1   305  and Q 2   310  may be bipolar PNP transistors. However, in accordance with the principles of the invention, these transistors are not so limited. Transistors Q 1   305  and Q 2   310  may be mismatched in the area of their respective emitters. In one embodiment, the ratio of emitter areas for Q 1   305  relative to Q 2   310  may be 5:6. The emitters of transistors Q 1   305  and Q 2   310  may be connected to the same rail. The bases of transistors Q 1   305  and Q 2   310  may be connected together. The collector of transistor Q 1   305  may be connected to one end of resistor R 1   315 , with the other end of resistor R 1  connected to current sources I 1   320 . A second resistor R 2   340  may be connected at one end to resistor R 1   315  and the collector of Q 1   305 , and at the other end to node  335 , the collector of Q 2   310 , the base of Q 3   330 , and current source I 2   325 . Resistor R 1   315  may have a value smaller than resistor R 2   340  of the embodiment of  FIG. 2 . In one embodiment, resistor R 1   315  may have a value of 200Ω, and resistor R 2   340  may have a value of 20 kΩ. 
   Current sources I 1   320  and I 2   325  may demand the same amount of current I from transistors Q 1   305  and Q 2   310 . Because transistors Q 1   305  and Q 2   310  are mismatched, Q 1   305  and Q 2   310  may provide different amounts of collector current. For the embodiment where the ratio of emitter areas for Q 1   305  and Q 2   310  is 5:6, 1/12 of current I may flow from the collector of Q 2   310  through resistor R 2   340 . A voltage drop equal to 1/12*I*R 2  may be produced across resistor R 2   340 . 
   As collector current flows from Q 1   305 , it will pass resistor R 1   315 , leading to the production of a second resistor voltage, or voltage drop, across resistor R 1   315  equal to I*R 1 . Using the circuit of  FIG. 3 , the base-emitter voltage V BEQ3  of transistor Q 3   330  may be approximately calculated as V BEQ3 ≈V BEQ1 −I*R 2 − 1/12*I*R 1 . The quiescent current I CQ3  of transistor Q 3  may be set and calculated as a function of the base-emitter voltage V BEQ3  of transistor Q 3  using the equation described below:
 
 I   CQ3   =e   ((−I*R2−1/12*I*R1)/26 mV) *16 *I   CQ1  
 
   For the embodiment where I=12 μA, R 1 =200Ω, and R 2 =20 kΩ, the quiescent current I CQ3  of transistor Q 3   330  may be 81 μA. The value 16 in the equation above may reflect the ratio between transistor Q 3   330  and transistor Q 1   305 . The embodiment of  FIG. 3  is capable of biasing and steering the quiescent current I CQ3  Of transistor Q 3   330  by changing the value of resistor R 1   315  or R 2   340 . 
     FIG. 4  is a schematic diagram of an embodiment of an output stage of an op-amp.  FIG. 4  may include the circuit embodiment of  FIG. 2  at the high side  485 , and a symmetrical circuit embodiment employing NPN transistors on the low side  490 . The symmetrical nature of the high side circuit  485  and the low side circuit  490  may produce a balanced op-amp, where I CQ3 =I CQ6 . The low side  490  may operate similarly to the embodiment of  FIG. 2 . For this embodiment, if transistor Q 4   455  has an emitter with an area ⅚th of the emitter of transistor Q 5   460 , and if current sources I 3   440  and I 4   445  demand equal amounts of current, then the base-emitter voltage for transistor Q 6   465  may be expressed as V BEQ6 =V BEQ5 − 1/12*I*R 3 , with I being the amount of current demanded by current sources I 3   440  and I 4   445 . 
   Certain embodiments disclosed herein describes a portion of an output stage of an op-amp. Any of these embodiments may be part of or connected at the various connection points shown in  FIGS. 1 through 3  to a known op-amp, H-bridge, or other circuit. The embodiments disclosed herein are not intended to be limited in use with any particular op-amp or other circuit. 
   Several features and aspects of the present invention have been illustrated and described in detail with reference to particular embodiments by way of example only, and not by way of limitation. Those of skill in the art will appreciate that alternative implementations and various modifications to the disclosed embodiments are within the scope and contemplation of the present disclosure. For example, the foregoing embodiments have been described using transistors of a particular type (e.g. n-type, p-type). It will be apparent that inputs and transistor types can be varied to as to vary the circuit configuration, while providing the same effect. Also, for example, the foregoing embodiments have been described with respect to bipolar transistors. It will be apparent that other transistors may be used instead, while providing the same effect. For instance, the present invention may be applicable to MOS transistors, with a gate of a MOS transistor corresponding to the base of a bipolar transistor, the drain of a MOS transistor corresponding to the collector of a bipolar transistor, and the source of a MOS transistor corresponding to the emitter of a bipolar transistor. Use of MOS transistors in place of bipolar transistors may eliminate any base current errors associated with bipolar transistors. Therefore, it is intended that the invention be considered as limited only by the scope of the appended claims.