Abstract:
Carrier signals are modulated by information (e.g., television) signals in a particular frequency range. The information signals are oversampled at a first frequency greater than any of the frequencies in the particular frequency range to provide digital signals at a second frequency. The digital signals are introduced to a carrier recovery loop which provides a feedback to regulate the frequency of the digital signals at the second frequency. The digital signals are introduced to a symbol recovery loop which provides a feedback to maintain the time for the production of the digital signals in the middle of the data signals. The gain of the digital signals is also regulated in a feedback loop. The digital signals are processed to recover the data in the data signals. By providing digital feedbacks, the information recovered from the digital signals can be quite precise. In one embodiment, the carrier signals are demodulated to produce baseband inphase and quadrature signals. The inphase and quadrature signals are then oversampled and regulated in the feedback loops as described above. In a second embodiment, the carrier signals downconverted to produce intermediate frequency signals which are oversampled to produce the digital signals at the second frequency without producing the inphase and quadrature signals. The oversampled signals are then regulated in the feedback loops as described above. In a third embodiment, the carrier signals are oversampled without being downconverted and without producing the inphase and quadrature signals.

Description:
This invention relates to a system for, and method of receiving information (e.g., video and/or data) signals transmitted by a satellite from a plurality of stations each operative in an individual frequency range and for recovering the information represented by the information signals. 
     BACKGROUND OF THE INVENTION 
     Satellites have been in existence for a number of years for receiving signals in space from a plurality of television stations and for transmitting these signals to a subscriber on the ground. Each of the television stations provides signals in an individual range of frequencies. For example, the encoded digital signals from the different television stations may have different data rates in a range between approximately two megabits/second (2 Mb/s) to approximately ninety megabits/second (90 Mb/s). 
     The satellites receive the signals from the different television stations in the frequency range of approximately 2-90 Mb/s and modulate these signals with a carrier signal having a suitable frequency such as a frequency in the range of approximately nine hundred and fifty megahertz (950 MHz) to approximately twenty one hundred and fifty megahertz (2150 MHz). The satellites then transmit the modulated carrier signals to television receivers on the ground. 
     The television receivers then convert the carrier signals to signals at an intermediate frequency such as approximately four hundred and eighty megahertz (480 MHz). These intermediate frequency signals are then demodulated at the television receivers and the demodulated signals are processed to recover the data signals from the individual ones of the television stations. The processing of the signals occurs on an analog basis. 
     It is well recognized that the processing of the signals on an analog basis to recover the data in the data signals is not as precise as would ordinarily be desired. The recovery of such data on a precise basis by analog techniques is especially difficult in view of the fact that the data signals may occur in a range of frequencies as great as approximately two megabits/second (2 Mb/s) to approximately ninety megabits/second (90 Mb/s). 
     BRIEF DESCRIPTION OF THE INVENTION 
     Carrier signals are modulated by information (video and/or data) signals in a particular frequency range. The information signals are oversampled at a first frequency greater than any of the frequencies in the particular frequency range to provide digital signals at a second frequency. 
     The digital signals are introduced to a carrier recovery loop which provides a feedback to regulate the frequency of the digital signals at the second frequency. The digital signals are also introduced to a symbol recovery loop which provides a feedback to maintain the time for the production of the digital signals in the middle of the information signals. The gain of the digital signals is also regulated in a feedback loop. The digital signals are processed to recover the data in the data signals. By providing digital feedbacks, the information recovered from the digital signals can be quite precise. 
     In one embodiment, the carrier signals are demodulated to produce baseband inphase and quadrature signals. The inphase and quadrature signals are then oversampled and regulated in the feedback loops as described above. 
     In a second embodiment, the carrier signals are downconverted to produce intermediate frequency signals which are oversampled to produce the digital signals at the second frequency without producing the inphase and quadrature signals. The oversampled signals are then regulated in the feedback loops as described above. 
     In a third embodiment, the carrier signals are oversampled without being downconverted or producing the inphase and quadrature signals. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     In the drawings: 
     FIG. 1 is a block diagram of a prior art receiver operative on an analog basis for receiving signals from a satellite and for recovering the information represented by such signals; 
     FIG. 2 is a block diagram of a receiver constituting one embodiment of the invention for receiving signals from a satellite and for processing such signals, primarily on a digital basis, to recover the information represented by such signals; 
     FIG. 3 is a block diagram of a receiver constituting a second embodiment of the invention, simplified in several respects relative to the embodiment shown in FIG. 2, for receiving signals from a satellite and for processing such signals, primarily on a digital basis, to recover the information represented by such signals; 
     FIG. 4 is a block diagram of a receiver constituting a third embodiment of the invention, simplified relative to the embodiments shown in FIGS. 2 and 3, for receiving signals from a satellite and for processing such signals, primarily on a digital basis, to recover the information represented by such signals; 
     FIG. 5 is a block diagram showing in additional detail certain of the stages included in the receiver represented by the block diagram of FIG. 2; 
     FIG. 6 is a block diagram showing in additional detail the same stages as are shown in FIG. 5 when such stages are modified for inclusion in the receivers represented by the block diagrams of FIGS. 3 and 4; 
     FIG. 7 is a circuit diagram showing in additional detail the construction of a complex multiplier shown in block form in FIG. 5; 
     FIG. 8 is a circuit diagram showing in additional detail the construction of a complex multiplier shown in block form in FIG. 6; 
     FIG. 9 is a circuit diagram in block form and shows in additional detail the construction of half band filters shown in block form in FIGS. 5 and 6; and 
     FIG. 10 provides voltage wave forms indicating how the circuitry shown in FIG. 4 operates to produce signals at a suitable frequency such as approximately thirty-two megahertz (32 MHz). 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 1 is a circuit diagram, primarily in block form, of a “Traditional Receiver Architecture” generally indicated at  10  and known in the prior art for use by a television subscriber for receiving signals from a satellite and for processing such signals to recover the information (e.g., video images) or data represented by such signals. The “Traditional Receiver Architecture” shown in FIG. 1 operates primarily on an analog basis to process the received signals and recover the information or data represented by such signals. 
     The system  10  shown in FIG. 1 includes a line  12  for receiving radio frequency (RF) carrier signals from a satellite (not shown) in a conventional manner. These carrier signals may have a suitable frequency such as nine hundred and fifty megahertz (950 MHz) or twenty one hundred and fifty megahertz (2150 MHz). The carrier signals received on the line  12  may be modulated by information (e.g., video and/or data) signals at a particular frequency in a frequency range such as approximately two megabits/second (2 Mb/s) to approximately ninety megabits/second (90 Mb/s). The particular frequency in this frequency range is dependent upon the particular television station which is being received by the subscriber at any instant. Only one (1) frequency is selected at any one time by the system shown in FIG. 1 for receiving data and processing such information. 
     The signals on the line  12  are introduced to a tuner  14  which is shown within broken lines in FIG.  1 . The tuner  14  includes a downconvert stage  16  and a surface acoustic wave filter (SAW)  18 . The stage  16  converts the signals at the carrier frequency to signals at an intermediate frequency such as approximately four hundred and eighty megahertz (480 MHz). The surface acoustic wave stage  18  constitutes a band pass filter which passes signals only to a particular frequency such as approximately four hundred and eighty megahertz (480 MHz). 
     The signals from the tuner  14  pass to an automatic gain control stage  20 . The signals from the automatic gain control stage  20  are in turn introduced to a pair of multipliers  22  and  24  which also respectively receive sine and cosine signals from a stage  26 . The operation of the stage  26  is controlled by a voltage controlled oscillator  28  having a center frequency at the intermediate carrier frequency of approximately 480 MHz. 
     The outputs of the multipliers  22  and  24  are respectively connected to low pass filters (LPF)  30  and  32 . Connections are respectively made from the filters  30  and  32  to analog-to-digital (A/D) converters  34  and  36 , the operations of which are controlled by the output from a voltage controlled oscillator  38 . The outputs from the converters  34  and  36  are introduced to the input of an automatic gain control loop  40 , the output of which controls the operation of the automatic gain control stage  20 . 
     The outputs of the converters  34  and  36  also respectively pass to filters  42  and  44  which may constitute suitable low pass filters such as Nyquist filters  42  and  44 . The outputs from the filters  42  and  44  are in turn introduced to a forward error correction (FEC) stage  46 , the output from which on a line  47  constitutes the information represented by the information signals modulating the carrier signals. 
     The outputs from the filters  42  and  44  are also introduced to stages  46  designated as a “Symbol Recovery Loop” and to stages  50  designated as a “Carrier Recovery Loop.” The output from the Symbol Recovery Loop  48  controls the operation of the voltage controlled oscillator  38  and the output from the Carrier Recovery Loop  50  controls the operation of the voltage controlled oscillator  28 . 
     The carrier signals modulated by the data signals are received on the line  12 . The modulated carrier signals are converted to an intermediate frequency (IF) of approximately 480 MHz by the tuner  14  and the IF signals are provided with a gain control as at  20 . Inphase and quadrature components of these IF signals are then respectively produced in the multipliers  22  and  24 . The carrier signals at the IF frequency are then removed from these signals at  30  and  32  so that only the information signals with the inphase and quadrature components remain. 
     The information signals passing from the filters  30  and  32  with the inphase and quadrature components are respectively converted to digital signals at a particular frequency in the converters  34  and  36 . The low frequency components of the digitized signals then respectively pass through the Nyquist filters  42  and  44 . Errors in the low frequency signals passing through the filters  42  and  44  are then corrected in the forward error correction stage  46 . The operation of the stage  46  in providing such corrections is known in the prior art. 
     The signals from the filters  42  and  44  may be considered to constitute baseband signals respectively including the inphase and quadrature components. These signals are introduced to the carrier recovery loop  50  which detects changes in the phases of such signals and produces voltage variations representing such phase changes. These voltage variations produce changes in the frequency of the signals from the voltage controlled oscillator  28 . Such changes in frequency in turn cause changes to occur in the frequencies of the inphase and quadrature signals in stage  26 . In this way, the operation of the stage  26  is regulated so that the sine and cosine signals from such stage coincide in frequency with the frequency of the signals from the stage  20 . 
     The baseband signals from the filters  42  and  44  are also introduced to the symbol recovery loop  48 . The loop  48  detects changes in the phases of these signals and produces voltage variations representing such phase changes. Such voltage variations produce changes in the frequency of the signals from the voltage controlled oscillator  38 . Such changes in frequency in turn cause changes to occur in the times at which the converters  34  and  36  operate to produce the digital signals. In this way, the analog-to-digital signals are produced in the middle of the times that the information signals are produced. This assures that the analog signals will be digitally sampled at the times when the analog signals represent valid information (e.g., data bits). 
     FIG. 2 shows, primarily in block form, a receiver generally indicated at  60  and constituting one embodiment of the invention. One primary way in which the embodiment shown in FIG. 2 differs from, and is superior to, the embodiment shown in FIG. 1 is that the embodiment shown in FIG. 2 provides digital feedback loops. Another primary way in which the embodiment shown in FIG. 2 differs from, and is superior to, the embodiment shown in FIG. 1 is that the embodiment shown in FIG. 2 is able to recover information from information signals in a frequency range as wide as approximately two megabits/second (2 Mb/s) to approximately ninety megabits/second (90 Mb/s). This cannot be accomplished by the analog system shown in FIG.  1 . 
     The embodiment shown in FIG. 2 includes the line  12 , the tuner  14  and the automatic gain control stage  20 . The signals from the stage  20  are introduced to the multipliers  22  and  24  as in the embodiment shown in FIG.  1 . The multipliers  22  and  24  also respectively receive sine and cosine signals from the stage  26 . However, the stage  26  receives signals at the IF frequency (such as approximately 480 MHz) from an intermediate frequency (IF) oscillator  62 . An advantage of the system shown in FIG. 2 is that the frequency of the signal from the oscillator  62  does not have to be precise. 
     The baseband signal from the multiplier  22  passes through the filter  30  which introduces the low frequency components of this signal to the analog-to-digital converter  34 . In like manner, the signal from the multiplier  24  passes through the filter  32  which introduces the low frequency components of this signal to the analog-to-digital converter  36 . The converters  34  and  36  are shown as being disposed within a broken rectangle  64 . The broken rectangle indicates an integrated circuit chip which applicant&#39;s assignee of record has designed and fabricated and which applicant&#39;s assignee of record has designated as the “BCM4200.” All of the stages within the rectangle  64  and on the BCM4200 chip are digital. 
     The signals from the filters  30  and  32  are in the data rate range of approximately two megabits/second (2 Mb/s) to approximately 90 megabits/second (90 Mb/s). The different frequencies in this range represent signals transmitted from different television stations and retransmitted by the satellite to the subscriber. A fixed oscillator  65  introduces free running signals to the converters  34  and  36  at a frequency at least twice the bandwidth of the information signals in the frequency range of approximately 1 megahertz (1 MHz) to approximately 45 megahertz (45 MHz). For example, the signals from the fixed oscillator  65  may be at a somewhat precise frequency such as approximately one hundred and twenty megahertz (120 MHz). This causes the oscillator  65  to oversample the information signals even at the highest frequency in such frequency range. In this way, the information signals are sampled several times in each cycle even at the highest frequency in the frequency range. 
     The signals from the converters  32  and  36  pass to a complex multiplier  66  the construction of which will be described in detail subsequently. The outputs from the complex multiplier  66  are in turn introduced to a variable interpolator  68 . Output connections are respectively made from the variable interpolator  68  to Nyquist filters  70  and  72  respectively corresponding to the filters  42  and  44  in FIG.  1 . The outputs from the filters  70  and  72  are connected to a forward error correction stage  74  corresponding to the stage  46  in FIG.  1 . 
     The outputs from the filters  70  and  72  are also introduced to a carrier recovery loop  76  and a symbol recovery loop  78 . Each of the loops  76  and  78  operates on a digital basis. The carrier recovery loop  76  may include a phase detector for detecting phase errors and may also include a loop filter. The output from the carrier recovery loop  76  passes to a direct digital frequency synthesizer (DDFS)  80  which may be a numerically controlled oscillator. The oscillator introduces sine and cosine signals to the complex multiplier  66 . 
     The symbol recovery loop  78  may be constructed in a manner similar to the construction of the carrier recovery loop  76  and may be considered to include a phase detector, a loop filter and a numerically controlled oscillator. A connection is made from the output of the symbol recovery loop  78  to the variable interpolator  68 . The outputs of the converters  34  and  36  are connected to an automatic gain control (AGC) loop  90  which introduces signals to the AGC stage  20  to regulate the gain of the analog signals at the IF frequency of 480 MHz. The AGC loop  90  operates on a digital basis. 
     The signals from the filters  70  and  72  are introduced to the carrier recovery loop  76  which detects changes in the phases of such signals and produces signals representing such changes in phase. These signals are filtered in the loop filter in the loop  76  and the filtered signals are introduced to the digital frequency synthesizer  80  to produce changes in the frequency of the signals from the synthesizer. Sine and cosine components of such signals are introduced from the synthesizer  80  to the complex multiplier  66  which combines these signals with the inphase and quadrature components of the digitized data signals from the converters  34  and  36 . In this way, the signals from the complex multiplier  66  are maintained at the frequency of the information signals even though the frequency of the signals from the oscillator  62  is not precise. 
     In like manner, the symbol recovery loop  78  detects changes in the phases of the signals from the filters  70  and  72  and produces signals representing such changes in phase. These signals are filtered in the loop filter in the loop  78  and the filtered signals are introduced to the digital frequency synthesizer in the loop  78  to produce changes in the frequency of the signals from the synthesizer. These signals cause the sampling of the digital signals to be provided in the middle of the period of time that each of the information signals is produced. 
     The AGC loop  90  operates digitally to regulate the gain of the signals from the tuner  14  at the intermediate frequency of approximately 480 MHz. As will be appreciated, a digital system is more precise than an analog system. This causes the variable interpolator  68  to provide an enhanced operation in the system shown in FIG.  2  and described above because there is essentially no variation in the gain of the signals from the AGC stage  20 . 
     Furthermore, the operation of the AGC loop  90  is enhanced because the signals introduced to the AGC loop have an error frequency. This error frequency results from the fact that the frequency of the signals from the IF oscillator  62  is not precise. As will be appreciated from the previous discussion, this error frequency is eliminated by the operation of the carrier recovery loop  76 . 
     The digital system shown in FIG. 2 has additional advantages over the analog system shown in FIG.  1 . The digital system shown in FIG. 2 is able to recover the information from information signals in a range of frequencies as low as approximately two megabits/second (2 Mb/s) and as high as approximately ninety megabits/second (90 Mb/s). This is accomplished in part by oversampling the analog signals from the filters  30  and  32  with the oscillator (e.g. at 120 MHz) at a frequency considerably greater than any of the frequencies in the range of approximately 1 MHz to approximately 45 MHz. 
     It is desirable that the complex multiplier  66  precede the variable interpolator  68 . This results in part from the fact that the IF oscillator  62  is not precise. For example, if the IF oscillator  62  provides an error such as approximately five megahertz (5 MHz) and the information signals have a bandwidth of approximately one megahertz (1 MHz), the complex multiplier  66  could not correct for the five megahertz (5 MHz) error if the complex multiplier  66  followed the variable interpolator  68 , since the sampling rate at the outputs of the variable interpolator would be approximately two megahertz (MHz). 
     FIG. 3 shows another embodiment, generally indicated at  100 , of the invention. This embodiment is similar to the embodiment shown in FIG. 2 in a number of respects. Because of this, like components or stages in FIG. 3 are given the same numerical indications as in the embodiment shown in FIG.  2 . However, in the embodiment shown in FIG. 3, a fixed oscillator  102  providing signals at a suitable frequency such as approximately four hundred and fifty megahertz (450 MHz) is connected to an input terminal of a multiplier  103 , another input terminal of which is connected to the automatic gain control stage  20 . The output from the multiplier  103  is accordingly at a frequency of approximately thirty megahertz (30 MHz). The output from the multiplier  103  is introduced to a low pass filter corresponding to the low pass filter  30  in FIG.  2 . 
     Another difference between the embodiments shown in FIGS. 2 and 3 is that a complex multiplier  106  corresponding in FIG. 3 to the complex multiplier  66  in FIG. 2 receives the output from an analog-to-digital converter  108  corresponding to the converter  34  in FIG. 2. A second input to the complex multiplier  106  in FIG. 3 constitutes a “0” signal on a line  109 . The “0” indication turns off one side of the complex multiplier  106  so that the inphase and quadrature components of the digitally converted signals are not provided to the complex multiplier, but rather the digitalized second intermediate frequency (IF 2 ) signal is provided to the complex multiplier. 
     The oscillator  110  preferably operates at a suitable frequency such as approximately one hundred and twenty megahertz (120 MHz). Since the signals introduced to the converter  108  are at a suitable frequency such as approximately thirty megahertz (30 MHz), the oscillator  110  oversamples on a 4:1 basis the signals introduced to the converter. 
     As will be seen, sine and cosine components are produced only at the outputs of the digital frequency synthesizer  80 . This is in the digital domain. Since the sine and cosine components are produced only in the digital domain, the down conversion from the IF frequency of 480 MHz to 30 MHz does not have to be precise. One reason is that the carrier recovery loop  76  provides precision in the frequencies provided to the complex multiplier  106 . In view of this, the frequency of the oscillator  110  does not have to be as precise as the frequency of the oscillator  38  in FIG.  1 . 
     FIG. 4 shows an embodiment which is even simpler in construction than the embodiment shown in FIG.  3 . In the embodiment of FIG. 4, the fixed oscillator  102 , the multiplier  103  and the low pass filter  104  shown in FIG. 3 are eliminated. Furthermore, a fixed oscillator  122  is provided with a suitable frequency such as 128 MHz and signals from this oscillator are introduced to an analog-to-digital converter  120  corresponding to the converter  108  in FIG.  2 . As a result, the signals at 480 MHz from the automatic gain control stage  20  are sampled at a frequency of approximately one hundred and twenty eight megahertz (128 MHz) in the analog-to-digital converter  120  which produces a digital signal at a second IF frequency of 32 MHz. As will be appreciated, the beat frequency of thirty-two (32 MHz) is obtained from the following relationship: 4(128)−480=32. This process is known as “sub-sampling.” 
     FIG. 10 illustrates how a beat frequency is obtained by introducing signal at 480 MHz and 128 MHz to the converter  120 . The signal at 480 MHz is illustrated schematically at  130  in FIG.  10 . The sampling at the frequency of 128 MHz causes signals to be produced at a frequency of 32 MHz. The signals at the frequency of 32 MHz are indicated schematically in FIG. 10 by dots  132 . 
     FIG. 5 illustrates in additional detail certain of the features in the system of FIG.  2 . The sub-system shown in FIG.  5  and generally indicated at  148  includes the analog-to-digital converters  34  and  36 , the complex multiplier  66 , the variable interpolator  68 , the carrier recovery loop  76 , the symbol recovery loop  78  and the digital frequency synthesizer  80  also shown in FIG.  2 . The converters  34  and  36  are shown as respectively receiving “I” and “Q” signals. The “I” and “Q” signals respectively indicate baseband inphase and quadrature signals. 
     An “and” network  150  is shown in FIG. 5 as having one input connected to the converter  36  and another input connected to receive a binary “1”. The binary “1” indicates that the output from the converter  36  is introduced to the complex multiplier  66 . Half band filters  152  and  154  are shown in FIG. 5 as being connected between the complex multiplier  66  and the variable interpolator  68 . As will be seen from the following discussion with respect to FIG. 9, each of the half band filters  152  and  154  divides the frequency range of 1-45 MHz into reduced frequency bands. One of these frequency bands is then selected in accordance with the individual one of the television channels selected for viewing by the subscriber. 
     The sub-system shown in FIG.  6  and generally indicated at  160  is intended to be used with the embodiments shown in FIGS. 3 and 4. The sub-system  160  in FIG. 6 is similar to the sub-system  148  in FIG. 5 except that the “and” gate  162  corresponding to the “and” gate  150  in FIG. 5 receives a logic “0” on one of its inputs. Because of this, the quadrature signal is not introduced to the complex multiplier  66 . 
     FIG. 9 illustrates the half band filters  152  and  154  in additional detail. One of the half band filters  152  and  154  is generally indicated at  170  in FIG.  9 . The other one of the half band filters  152  and  154  is constructed in a similar manner. In FIG. 9, a line  172  is provided to receive the signals from the complex multiplier  68 . A plurality of half band filters  174 ,  176 ,  178 ,  180  and  182  are connected in series with the line  172  and with one another. The output from the line  172  and from the filters  174 ,  176 ,  178 ,  180  and  182  is connected to a 6:1 multiplexer  184 . 
     The line  172  and each of the half band filters pass information signals at an individual range of symbol rates. Each of the filters  174 ,  176 ,  178 ,  180  and  182  passes signals at symbol rates one half of the rate introduced to the previous filters in the chain. This may be seen from the following table: 
     
       
         
               
               
               
             
           
               
                   
                   
               
               
                   
                 Output 
                 Symbol Rate in Megabaud 
               
               
                   
                   
               
             
             
               
                   
                 Line 172 
                 22.5-45.0 
               
               
                   
                 Filter 174 
                 11.25-22.5  
               
               
                   
                 Filter 176 
                 5.625-11.25 
               
               
                   
                 Filter 178 
                 2.8125-5.625  
               
               
                   
                 Filter 180 
                 1.40625-2.8125  
               
               
                   
                 Filter 182 
                 0.703125-1.40625  
               
               
                   
                   
               
             
          
         
       
     
     The output from only one of the line  172  and the filters  174 ,  176 ,  178 ,  180  and  182  can pass through the multiplexer  184  at any instant. 
     FIG. 7 shows in additional detail the construction of the complex multiplier  66  in the embodiment shown in FIG.  2 . In the embodiment shown in FIG. 7, the outputs from the converters  34  and  36  are respectively shown on lines  200  and  202 . The output on the line  200  is introduced to multipliers  204  and  208  and the output on the line  202  is introduced to multipliers  206  and  210 . 
     The multipliers  204  and  210  receive a second input from an output line  212  from the digital frequency synthesizer  80  and the multipliers  206  and  208  receive a second input from an output line  214  from the digital frequency synthesizer  80 . The inputs to the multipliers  204  and  210  represent a cosine function and the inputs to the multipliers  206  and  208  represent a sine function. 
     Connections are made from the outputs of multipliers  204  and  206  to a subtracter  212 . The output from the subtracter  212  is introduced through a line  214  to the half band filter  152  in FIGS. 5 and 6. In like manner, the outputs from the multipliers  208  and  210  are introduced to an adder  216  in FIG.  7 . The output from the adder  216  passes through a line  218  to the half band filter  154  in FIGS. 5 and 6. 
     FIG. 8 shows in additional detail the complex multiplier  66  in FIGS. 3 and 4. As will be appreciated from the showing in FIGS. 5 and 6 and from the above discussion, the Q output on the line  202  is zero. This is represented by the introduction of a “0” to the subtracter  212  and adder  216 . In this way, the embodiment shown in FIG. 8 does not provide inphase and quadrature functions. 
     A variable interpolator for use as the variable interpolator  68  is known in the prior art. The variable interpolator  68  may be constructed in accordance with the disclosures of any of the following publications: 
     Gardner, Floyd M., “Interpolation in Digital Modems—Part I: Fundamentals”, IEEE Transactions on Communications, No. 3, March 1993. 
     Harris, Fred. “On the Relationship Between Multirate Polyphase FIR Filters and Windowed, Overlapped, FFT Processing”, Proceedings of the Twenty-Third Asilomar Conference on Signals, Systems and Computers, Oct. 30-Nov. 1, 1989. 
     Harris, Fred, et al. “Modified Polyphase Filter Structure for computing Interpolated Information As Successive Differential Corrections”, Proceedings of the 1991 International Symposium on Circuits and Systems, Singapore, Jun. 11-14, 1991. 
     Crochiere, Ronald E. and Rabiner, Laurence R., Multirate Digital Signal Processing; Englewood Cliffs, N.J.: Prentice Hall 1983. 
     U.S. Pat. No. 5,504,785—Apr. 2, 1996—Digital Receiver for Variable Symbol Rate Communication, Inventors: Donald W. Becker, Fred Harris, James C. Tiernan. 
     Although this invention has been disclosed and illustrated with reference to particular embodiments, the principles involved are susceptible for use in numerous other embodiments which will be apparent to persons of ordinary skill in the art. The invention is, therefore, to be limited only as indicated by the scope of the appended claims.