Abstract:
A method for digital phase detection, comprises the steps of: providing a reference clock; receiving a feedback clock; determining a timing difference between the reference clock and the feedback clock; determining a polarity that indicates the leading or lagging relationship between the reference clock and the feedback clock; adaptively selecting one of at least two operating modes for generating a quantized level indicative of the timing difference, wherein in a first operating mode the quantized level is a constant maximum value and wherein in a second operating mode the quantized level is proportional to the timing difference; and generating a digital phase detection output as a combination of the polarity and the quantized level.

Description:
FIELD OF INVENTION 
     This invention relates to a digital phase and frequency detector (“DPFD”), and, in particular, to a digital phase and frequency detector that can separately detect the polarity and magnitude of the phase difference of input signals and has at least two operating modes which can be adaptively switched. 
     BACKGROUND 
     Generally in electronics, a digital phase and frequency detector (“DPFD”) is a device which compares the phase and frequency of two input signals. The two inputs usually correspond to a voltage-controlled oscillator (“VCO”), as a feedback signal, and an external source as a reference signal. The DPFD can have two outputs for providing information to other circuitry for phase locking. To form a phase-locked loop (“PLL”), the DPFD&#39;s phase error output is fed to a loop filter which integrates the signal to smooth it. This smoothed signal is fed to a voltage-controlled oscillator which generates an output signal with a frequency that is proportional to the input voltage. The VCO output is also fed back to the DPFD to form the PLL circuit. 
     In traditional analog PLL techniques, a phase and frequency detector converts the phase and frequency differences between the reference clock and the feedback clock into an analog signal. Depending on the technology, this signal can either be a voltage which can be applied directly to the loop filter or a current generated by a charge pump. One disadvantage of the analog PLL structures is the silicon integration. Due to spur reduction requirements, the loop filter requires large resistors and capacitors, typically connected externally to an IC chip, to achieve a low PLL bandwidth. Another major disadvantage of conventional PLLs is lack of portability from one process technology to another. Therefore, more and more PLL designers are attempting to use digital architectures to implement a PLL. One of the key function blocks of a digital PLL is a digital phase and frequency detector. 
     Typically, a digital phase and frequency detector is implemented based on a time-to-digital converter (“TDC”). Various approaches for digital phase and frequency detectors have been attempted but have draw backs, such as increased hardware resources, slow locking time, or degradation in performance. For instance, it is difficult to achieve a DPFD that has both accuracy and low complexity. Furthermore, a bang-bang PLL technique is simpler since it uses a one-bit polarity output DPFD, but is not accurate in high performance applications. Thus, the need exists for a digital phase and frequency detector that can reliably produce phase error signals and has relatively low complexity. 
     SUMMARY OF INVENTION 
     An object of this invention is to provide a digital phase and frequency detector that has at least two operating modes to optimize hardware resources, locking time, and power consumption based on the phase error detection. 
     Another object of this invention is to provide a digital phase and frequency detector that can adaptively switch between operating modes depending on a number of polarity changes of the phase error. 
     Yet another advantage of this invention is to provide a digital phase and frequency detector that can increase the resolution of the phase detection while minimizing the hardware resources to determine the phase error. 
     Briefly, the present invention relates to a method for digital phase detection, comprising the steps of: providing a reference clock; receiving a feedback clock; determining a timing difference between the reference clock and the feedback clock; determining a polarity that indicates the leading or lagging relationship between the reference clock and the feedback clock; selecting one of at least two operating modes for generating a quantized level indicative of the timing difference, wherein in a first operating mode the quantized level has only a polarity information and wherein in a second operating mode the quantized level has both a polarity information and a magnitude information, where the magnitude information is proportional to the input phase error. 
     An advantage of this invention is that a digital phase and frequency detector is provided that has at least two operating modes to optimize hardware resources, locking time, and power consumption based on the phase error detection. 
     Another advantage of this invention is that a digital phase and frequency detector is provided that can switch between operating modes depending on a number of polarity changes of the phase error. 
     Yet another advantage of the present invention is that a digital phase and frequency detector is provided that can increase the resolution of the phase detection while minimizing the hardware resources to determine the phase error. 
    
    
     
       DESCRIPTION OF THE DRAWINGS 
       The foregoing and other objects, aspects, and advantages of the invention will be better understood from the following detailed description of the preferred embodiment of the invention when taken in conjunction with the accompanying drawings in which: 
         FIG. 1  illustrates a transfer curve for a digital phase and frequency detector of the present invention. 
         FIG. 2  illustrates a schematic for a digital phase and frequency detector of the present invention having a transfer curve with two operating modes. 
         FIG. 3  illustrates a schematic for a clock edge interval generation circuit of the present invention. 
         FIG. 4  illustrates a waveform diagram for various signals received and outputted by a clock edge interval generation circuit of the present invention. 
         FIG. 5  illustrates a Vernier-Delay line time-to-digital converter of the present invention. 
         FIG. 6  illustrates a schematic for a locking detector of the present invention to determine whether a “locking” state is met. 
         FIG. 7  illustrates a waveform diagram for various signals of a digital phase and frequency detector of the present invention. 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       FIG. 1  illustrates a transfer curve for a digital phase and frequency detector of the present invention. A digital phase and frequency detector (“DPFD”) of the present invention can have at least two operating modes, including a linear operating mode and a saturation operating mode. In the linear operating mode, the DPFD outputs a quantized level to represent both the polarity and magnitude of the input phase error, Φ E , between a first input signal, e.g. a reference clock, and a second input signal, e.g. a feedback clock. In the saturation operating mode, the DPFD outputs a maximum constant level to represent the polarity of the input phase error between the first input signal and the second input signal. A transfer curve can be used to represent the quantized levels of the digital phase and frequency detector as a function of the input phase error of the two signals. Furthermore, the transfer curve can have a saturation region where the DPFD is in a saturation operating mode and a linear region where the DFPD is in a linear operating mode. 
     If the input phase error ranges from a negative threshold value, −TH, to a positive threshold value, +TH, inclusively, then the input phase error can be converted to a quantized level according to a linear formula, e.g.,
 
 DPFD   out   =k   DPFD ×( T   error   /t   LSB ),  (1)
 
where k DPFD  can be a value set by a user, t LSB  is the delay of a buffer in the buffer chain, T error  can be converted from the phase error and is given by the formula below:
 
 T   error =(φ phase     —     error /2π)× T   reference ,  (2)
 
where the T reference  is a period of the reference clock. Depending on the application, the number of quantized levels can preferably range from 32 to 128 distinct levels for the output of the DPD.
 
     When the input phase error is smaller than the negative threshold value, then a maximum negative quantized level is outputted to reflect this difference. When the phase difference is greater than the positive threshold value, then a maximum positive quantized level is outputted to reflect this difference. These two cases can be referred to as the saturation region on the transfer curve. The input phase error can be said to be in a saturation region, when the absolute value of input phase error falls in this region. 
     To aid in the understanding of the present invention, the following disclosed circuits relate to a DPFD having a saturation operating mode and a linear operating mode. However, it is understood that other equivalent circuits may be used to implement a DPFD having at least two operating modes. 
       FIG. 2  illustrates a schematic for a DPFD of the present invention having two operating modes. The DPFD comprises a phase error magnitude detector  2 , a phase error polarity detector  4 , and a locking detector  6 . 
     The phase error magnitude detector  2  measures the magnitude of the phase difference between the reference clock and the feed-back clock. The phase error magnitude detector  2  comprises a clock edge interval generation circuit and a Vernier-Delay line time to digital converter (“VD-TDC”)  10 . The clock edge interval generation circuit generates the start and stop signals to be inputted to the VD-TDC  10 . The clock edge interval generation circuit can generate a clock edge interval which is equal to the rising edge interval between the reference clock and the feed-back clock. Based on the start and stop signals, the phase error magnitude detector  2  can measure the input phase error. 
     The phase error polarity detector  4  determines whether the reference clock is leading or the feed-back clock is leading, and outputs the respective polarity to indicate that relationship. For instance, if the reference clock is leading the feed-back clock, then a positive polarity (e.g., a high value or a “1” value) may be outputted. Conversely if the reference clock is lagging the feed back clock, then a negative polarity (e.g., a low value or a “0” value) may be outputted. 
     The determined polarity and the calculated input phase error can be multiplied and outputted by the DPFD. The locking detector  6  can determine when a locking state has occurred by counting the number of polarity changes. If the number of polarity changes has reached a predefined number, then a locking flag is activated to indicate the locking state to the VD-TDC  10 . The locking flag can be deactivated when the target frequency changes or a reset signal is received. 
     If the locking flag is activated, then the DPFD is operated in the linear operating mode, where the VD-TDC  10  outputs a quantized level corresponding to the input phase error. If the locking flag is not activated, then the DPFD is operated in the saturation operating mode, where the VD-TDC  10  outputs the maximum quantized level. 
       FIG. 3  illustrates a schematic for a clock edge interval generation circuit. The clock edge interval generation circuit comprises a first data flip flop DFF 1 , a second data flip flop DFF 2 , a logic OR gate  12  and a logic AND gate  14 , and a delay circuit  16 . The delay circuit  16  can be used to overcome a “dead zone” problem. The first data type flip flop DFF 1  receives the reference clock and a high signal (i.e., “1”) on its input and outputs a signal A. The second data type flip flop DFF 2  receives the feed-back clock and a high signal (i.e., “1”) on its input and outputs a signal B. The signal A and signal B are applied to the logic OR gate  12 . The logic OR gate  12  outputs the result (i.e., A+B) to the start input of the VD-TDC  10 . The signal A and signal B are applied to the logic AND gate  14 . The logic AND gate  14  outputs the result (i.e., A&amp;B) to the stop input of the VD-TDC  10 . 
       FIG. 4  illustrates a waveform diagram for the various signals received and outputted by the clock edge interval generation circuit. The signals A and B follow the reference clock and feed-back clock, respectively. The signal A+B is high when one of the signals A or B is high. The signal A&amp;B is high when both A and B are high. The input phase error difference is given by the intervals 1-8 which illustrate the difference in phase between the reference clock and the feed-back clock for a given time interval. 
       FIG. 5  illustrates a VD-TDC of the present invention having a slow buffer chain and a fast buffer chain. The VD-TDC comprises a slow buffer chain  50 , a fast buffer chain  52 , an overflow circuit, and a TDC decoder, where each of the buffer chains has L buffers. The delay of a single buffer of the slow buffer chain is T 1  where the counterpart of the fast buffer chain is T 2 . T 1  must be larger than T 2  to guarantee a proper operation of the VD-TDC. The resolution of the VD-TDC is determined by the difference between T 1  and T 2 . The non-saturation range of the VD-TDC depends both on the T 1  and T 2  difference and the number of the delay stages in a chain. 
     The delay of a buffer in the slow chain is slightly greater than the delay of a buffer in the fast chain. As the start and stop signals propagate to their respective delay chains, the timing difference between the buffers is decreased in each Vernier stage by the difference in the buffer delays. 
     In each stage, the signals are fed into an arbiter circuit (e.g., a DFF in the VD-TDC of the present invention), which decides which of the two pulses is leading. The position in the delay line at which the stop signal catches up with the start signal gives information about the measured time between the start and stop signals. 
     Depending on the application, the resolution (which can be equal to the difference in buffer delays) is made less than several picoseconds and the non-saturation range of the VD-TDC is made less than 2 to 4 digital controlled oscillator (“DCO”) periods, e.g., the highest frequency generator in a DPLL. When the time interval between the start and stop signal is larger than the non-saturation range of the VD-TDC, the VD-TDC will generate an overflow signal and the output of the VD-TDC will be a constant maximum value. 
     Referring to  FIG. 2 , the phase error polarity detector  4  comprises a data type flip-flop DFF 3  and a polarity converter. It is to be understood that the phase error polarity detector  4  can be implemented by other equivalent circuits for polarity detection. When the signal B has a rising edge, the DFF 3  can sample the valve of the signal A. If the sampled value is high, the polarity converter will output a positive sign flag. Alternatively, if the sampled value is low, the polarity converter will output a negative sign flag. Once the polarity of the signal is determined, the output of the phase error magnitude detector  2  and the phase error polarity detector  4  are combined (e.g., multiplied together) to generate the DPFD output signal. 
       FIG. 6  illustrates a block diagram for a locking detector of the present invention. The structure of the locking detector comprises a clock dividing circuit  60 , a cyclic reset counter  62 , and a comparator  64 . When a PLL is close to a locking state, the polarity of the input phase error will hop between positive and negative polarities. The cyclic reset counter  62  can count the number of the polarity transitions. Every predefined number of feed-back clock cycles (e.g., 8), the comparator  64  will compare the output value of the counter  62  with a window value, where the window value can be one or more criteria set by a user to judge whether the loop is locked or not. If the window value is less than the counted number, the locking flag will change from a low signal to a high signal to predict the PLL is close to a locking state. 
     At the initial state when the PLL is not locking, the input phase error (or the distance between the rising edges of the reference clock and the feed-back clock) at the input of the DPFD is much larger than the non-saturation range of the VD-TDC. Thus, the VD-TDC can be operating in the saturation operating mode. Under this condition, the locking flag of the locking detector will be a low signal, which will turn off the buffer chains of the VD-TDC to save power. Thus, when the VD-TDC is off, the output of the VD-TDC can be treated as overflow. In the saturation operating mode, the decoder of the VD-TDC will output a constant maximum level until the locking flag goes to a high signal. 
     In addition to phase detection, the DPFD can also detect frequency errors between the reference clock and the feedback clock.  FIG. 7  illustrates a waveform diagram for various signals of a digital phase and frequency detector of the present invention, where the frequency differs between the reference clock and the feedback clock. In this example, the reference clock can be at 40 Mhz with a 25 ns initial delay and the feed back clock can be at 80 Mhz with a 0 ns initial delay. 
     When the reference clock is much slower than the feedback clock, signal B will be reset after a small amount of time when signal A goes high. The rising edge of signal A is lagging the rising edge of the signal B. Furthermore, the output of the phase error polarity detector (e.g., DFF 3 ) is low, which means a negative polarity. Thus, this kind of DPFD output corresponds to a negative frequency error between the reference clock and the feedback clock. Therefore, the DPFD can adjust the frequency of the feedback clock accordingly to compensate for the negative frequency error. 
     Conversely, when the reference clock is much faster than the feedback clock, the output of the phase error polarity detector (e.g., DFF 3 ) will always be high, which corresponds to a positive frequency error between the reference clock and the feedback clock. Thus, the DPFD can also adjust the frequency of the feedback clock accordingly to compensate for the positive frequency error. 
     While the present invention has been described with reference to certain preferred embodiments or methods, it is to be understood that the present invention is not limited to such specific embodiments or methods. Rather, it is the inventor&#39;s contention that the invention be understood and construed in its broadest meaning as reflected by the following claims. Thus, these claims are to be understood as incorporating not only the preferred methods described herein but all those other and further alterations and modifications as would be apparent to those of ordinary skilled in the art.