Abstract:
The apparatus includes a series active continuous time voltage regulator operating in conjunction with a alternating current power source and one or more loads. The alternating current power source is a voltage source that induces currents at a first end of the apparatus. At a second end of the apparatus one or more loads consume power from the apparatus. The series buck-boost regulator is composed of a pure monochromatic voltage source of frequency equal to that of the alternating current power source, and of constant phase with respect to the alternating current power source. The regulator is further composed of a sampling network that provides a scaled continuous time sample of the voltage delivered by the power conditioner to the loads. Finally, the regulator is composed of a high gain differential amplifier. The components of the regulator are configured to operate as a continuous time feedback control system that generates an error correction voltage that, when added to the voltage of the alternating current power source, results in an output voltage from the power conditioner that is a scaled replica of the monochromatic reference voltage. As a result, the voltage delivered by the power conditioner to the loads is substantially corrected of spectral impurities and impervious to the specific conditions of the alternating current power source or of the loads.

Description:
CROSS REFERENCE TO RELATED APPLICATIONS  
       [0001]    This application is a continuation of co-pending U.S. Ser. No. 60/267,399, filed on Feb. 7, 2001. The priority of the prior application is expressly claimed and its disclosure is hereby incorporated by reference in its entirety. 
     
    
     
       FIELD OF THE INVENTION  
         [0002]    This invention relates generally to power conditioners and methods which operate in the presence of unconditioned alternating current power, and more particularly to series active regulators used as active filters. These systems act to remove spectral impurities from the alternating current power line so as to provide power that is substantially sinusoidal and devoid of harmonic, spurious and random noise components.  
         BACKGROUND OF THE INVENTION  
         [0003]    Power conditioning circuits are used to remove unwanted voltages and currents from alternating current voltage sources intended for supplying operating power to electronic equipment. Power conditioning sometimes refers to surge and spike protection. Another example of power conditioning pertains to removal of unwanted spectral components that may be of small amplitude with respect to the fundamental voltage of the power source. Specifically, power conditioning may refer to removal of spectral components either introduced by the power source or generated by the loads. Finally, power conditioning sometimes refers to regulation of the average amplitude of the power source voltage.  
           [0004]    A series regulator is an apparatus that acts to control the voltage at some node in a network by adding a correction voltage to an existing voltage so that the sum of the voltages is controlled. A source of correction (or error correction) voltage is connected in series with an existing source of voltage. The correction voltage either acts to buck or to boost the existing voltage so that the discrepancy between the existing voltage and the desired voltage is removed. Voltage “bucking” is the addition of a opposite polarity voltage to an existing voltage, the result of which is a new voltage that has reduced magnitude with respect to the existing voltage. Voltage “boosting” is the addition of a same polarity voltage to an existing voltage, the result of which is a new voltage that has increased magnitude with respect to the existing voltage.  
           [0005]    A feedback control system is a type of system that actively minimizes the error formulated as the difference between its desired behavior and its actual behavior. A feedback control system makes continuous comparison between the actual behavior of a system and a standard of desired behavior. From this comparison, a correction influence is derived that acts on the system to null the error.  
           [0006]    An inverter is an apparatus that converts DC voltage into AC voltage. It may also be configured to first convert AC voltage into DC voltage, and second to convert DC voltage back into AC voltage. Inverters are frequently used in power conditioning and power control circuits to perform corrective operations on the incoming AC power.  
           [0007]    A growing number of nonlinear loads in the electric utility power network has resulted in increasing waveform distortion of both voltages and currents in ac power distribution systems. Typical nonlinear loads are computer controlled data processing equipment, numerical controlled machines, variable speed motor drives, robotics, medical and communication equipment.  
           [0008]    Utilities provide sinusoidal supply voltages. Nonlinear loads draw square wave or pulse-like discontinuous currents instead of the purely sinusoidal currents drawn by conventional linear loads. As a result, nonlinear current flows through the predominantly inductive source impedance of the electric supply network. Consequently, a non-linear load causes load current harmonics and reactive power to flow back into the power source. This results in unacceptable voltage harmonics and supply load interaction in the electric power distribution network.  
           [0009]    The degree of current or voltage distortion can be expressed in terms of the magnitudes of harmonics in the waveforms relative to the fundamental magnitude. Total Harmonic Distortion (THD) is one of the accepted standards for measuring voltage or current quality in the electric power industry.  
           [0010]    Aside from the waveform distortion, another imperfection common to utility electric power is the presence of spurious and random noise. Various electric appliances such as electric motors, radio and television receivers, and digital electronic computers and related appliances generate noise components that are non-harmonically related to the fundamental power line frequency. Finite isolation between the appliances and the public utility power grid results in spurious and random noise energy components traveling along the power grid. Further, electromagnetic radiation containing a wide variety of spectral components as well as broad spectrum noise couples into the power grid and propagates to the end user load equipment.  
           [0011]    A wide variety of electrically powered appliances such as comprises home entertainment audio and video systems, test and measurement systems, hospital monitoring equipment, and computer systems have degraded operating performance when powered by AC power that contains typical levels of harmonic, spurious, and random noise components.  
           [0012]    Another problem related to the nature of utility AC power pertains to the finite impedance of the AC power source as it appears at the load. When a number of separate pieces of electrical equipment are connected to the same AC power source there is a finite degree of coupling between the components at the point of common connection to the power source. The noise that each component generates and that leaks over to the AC input is allowed to enter each of the other components in the system. In this way each part of the system operates to degrade the performance of the remaining components. The AC power source offers only partial isolation between components because it has a finite source impedance at the point of common connection to the loads. An example of this is digital clock noise generated by a digital to analog converter in an audio playback system coupling through the power line connections and degrading the performance of a preamplifier in the same system.  
           [0013]    Finite source impedance also inhibits the performance of electrical equipment by allowing the AC voltage sinewave to sag during high current transients. This results in higher levels of power supply ripple and compromised performance. An example can be seen in an audio amplifier application, where signal transients require high levels of current with sustained power supply voltage. In this case the finite power line impedance results in temporary collapsing of the AC sinewave voltage and a resulting distortion from the amplifier.  
           [0014]    The unbalanced configuration of single phase AC power introduces additional problems for sensitive electronic equipment. There is significant radiation and coupling of AC power energy into downstream circuitry because the fields associated with unbalanced power are far reaching. In addition, the AC return current path utilizes the neutral line, which is eventually tied to earth ground. Since earth ground is the common node for each component in a system a certain degree of AC noise injection into the rest of the system is made possible by terminating the AC current path to this node.  
           [0015]    A number of techniques have been used in an attempt to provide power line conditioning to address some of the foregoing problematic conditions. Passive filters, such as LC tuned filters, are often used because they are efficient and inexpensive. There are, however, a number of problems associated with passive filters. Practical constraints of size and cost limit passive filters to relatively simple topologies using relatively few components. The result is only a moderate suppression of AC power imperfections. In addition, the tuned resonators in passive filters have non-ideal behavior at frequencies other than the power line frequency. This manifests itself as undesirably high source impedance and results in poor current on demand characteristics and poor inter-component isolation.  
           [0016]    Active power filters have been developed to resolve some of the problems associated with passive filters. Active power filters, or active power line conditioners (APLCs), inject signals into an ac system to cancel harmonics.  
           [0017]    Active filters comprise one or two pulse width modulated inverters in a series, parallel, or series-parallel configuration (with respect to the load or supply). The inverters have a dc link, which can be a dc inductor (current link) or a dc capacitor (voltage link). It is necessary to keep the energy stored in the dc link (capacitor voltage or inductor current) at an essentially constant value. The voltage on the dc link capacitor can be regulated by injecting a small amount of real power into the dc link. The injected real power compensates for the switching and conduction losses inside the APLC.  
           [0018]    One problem associated with active filters is that it is expensive and complicated to generate a reference signal for the feedback control that is sufficiently monochromatic and free of noise, and also phase locked to the incoming AC power. Customarily the digital clocks and phase detectors used to synchronize a free running oscillator to the AC line voltage generate considerable noise. The low loop bandwidth required for operation at power line frequencies results in inordinately long settling times and poor transient response.  
           [0019]    Another problem with active filters is that of stability. The feedback control loop acting to monitor and correct error in the output signal with respect to the reference is typically band limited to prevent oscillation of the loop. The spectral purity and desirably low output impedance of the output power is limited to the bandwidth of the loop.  
           [0020]    Alternatively, the AC power can be completely regenerated from a DC power supply operating from the AC power line. In this approach a pure sinewave source is amplified to the line voltage level by class A-B amplifiers. This approach suffers from very low efficiency, theoretically not to exceed 50%, and usually much lower values are realized due to practical circumstances. In addition, the source oscillator is typically derived from a digitally sampled sinewave and is corrupted by quantization impurities when reconstructed.  
         SUMMARY OF THE INVENTION  
         [0021]    A power line conditioner operated in a method according to the present invention utilizes a series buck-boost voltage regulator as a filter for removing unwanted impurities from a source of alternating current power. The preferred embodiment of the apparatus includes a transformer and an active filter coupled in series to a power distribution network. The power distribution network includes a voltage source that induces input currents at a first end of the power distribution network. Nonlinear loads and other conditions on the power distribution network cause harmonic, spurious, and random noise components to corrupt the power signals. The active filter of the invention uses a monochromatic reference derived from the incoming AC power voltage which is prepared by stripping off unwanted harmonics and noise and regulating amplitude to a predetermined value. A feedback control system is configured to operate in series with a secondary winding of the transformer so as to effectively subtract voltage imperfections from the incoming AC. The control loop is compensated in such a way that it is stabilized without compromising bandwidth. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0022]    For a better understanding of the nature and advantages of the invention, reference should be made to the following detailed description taken in conjunction with the accompanying drawings, in which:  
         [0023]    FIGS.  1 - 2  depict an embodiment of the feedback control loop of the present invention operating with an external power source.  
         [0024]    [0024]FIG. 3 depicts an embodiment of a pair of feedback control loops operating with an isolation transformer to form a balanced output power conditioner.  
         [0025]    [0025]FIG. 4 represents an embodiment of a single feedback control loop operating with an isolation transformer to form an unbalanced output power conditioner.  
         [0026]    [0026]FIG. 5 illustrates an embodiment of a voltage sampler of the present invention.  
         [0027]    [0027]FIG. 6 depict a first embodiment of the amplifiers of the present invention.  
         [0028]    [0028]FIG. 7 depicts a second embodiment of the amplifiers of the present invention.  
         [0029]    [0029]FIG. 8 depicts a third embodiment of the amplifiers of the present invention.  
         [0030]    [0030]FIG. 9 illustrates an embodiment of a conventional power regeneration system.  
         [0031]    [0031]FIG. 10 depicts an exemplary voltage reference according to one embodiment of the present invention.  
         [0032]    [0032]FIG. 11 depicts an exemplary voltage reference according to a second embodiment of the present invention.  
         [0033]    [0033]FIG. 12 illustrates an embodiment of a voltage reference circuit including voltage sampling means.  
         [0034]    [0034]FIG. 13 shows a sampling comparator of a preferred embodiment.  
         [0035]    [0035]FIG. 14 shows an embodiment of an alternative comparator.  
         [0036]    [0036]FIG. 15 shows an active low pass filter according to a preferred embodiment.  
         [0037]    [0037]FIG. 16 depicts an embodiment of a passive filter used in a preferred embodiment.  
         [0038]    [0038]FIG. 17 illustrates an embodiment of a phase splitter used in a preferred embodiment.  
         [0039]    [0039]FIG. 18 shows an embodiment of the present invention wherein amplifier supply voltage and auxiliary supply voltage are generated by the isolation transformer.  
         [0040]    [0040]FIG. 19 shows another embodiment of the present invention wherein amplifier supply voltage and auxiliary supply voltage are generated by the isolation transformer.  
         [0041]    [0041]FIG. 20 shows an embodiment of a first R-C network and its implementation for the purpose of stabilizing the feedback control loop.  
         [0042]    [0042]FIG. 21 shows an embodiment of a second R-C network and its implementation for the purpose of stabilizing the feedback control loop.  
         [0043]    [0043]FIG. 22 shows an embodiment of a third R-C network and its implementation for the purpose of stabilizing the feedback control loop.  
         [0044]    [0044]FIG. 23 shows an embodiment of a fourth R-C network and its implementation for the purpose of stabilizing the feedback control loop.  
         [0045]    [0045]FIG. 24 shows an embodiment of a fifth R-C network and its implementation for the purpose of stabilizing the feedback control loop. 
     
    
     DETAILED DESCRIPTION OF THE INVENTION  
       [0046]    In the description that follows, like reference numerals refer to corresponding parts throughout the several views of the drawings.  
         [0047]    [0047]FIG. 1 depicts elements associated with an embodiment of the present invention. In general, the invention is directed to a series configured error correction feedback control system  100  which operates with other familiar components and a source of alternating current power  200 . The feedback control system  100  is composed of sampler  11 , voltage reference  12 , and amplifier  13 . All voltages describing the behavior of control system  100  are taken with respect to output  502 , which is typically tied to system ground. It is a general characteristic of the invention that sampled voltage is compared with a reference voltage by an amplifier. It is a further characteristic of the invention that the sampled voltage appears at an input of an amplifier as negative feedback, whereby the amplifier acts in series with an alternating current power source to provide an output voltage as a scaled but otherwise substantially identical voltage to the reference voltage. An advantage of this embodiment of the system is to provide harmonic, spurious, and random noise isolation between source and load over a multi-decade frequency range (e.g., 100 Hz to 100 Khz). Another advantage of this embodiment of the system is to provide fractional ohm source impedance (e.g., 0.01 ohm to 0.1 ohm) to the load over a multi-decade frequency range.  
         [0048]    It is ordinarily desirable that the reference voltage  15  be a pure sinewave. It is likewise ordinarily deemed desirable that the voltage output  501  of the power conditioner be as nearly perfect a sinewave as possible because the power supplies in electronic equipment are not well suited to remove broad band spectral impurities. Should something other than a perfect sinewave be desired for special purposes, the reference voltage  15  of voltage reference  12  must be correspondingly non-sinusoidal.  
         [0049]    It is the normal operation of feedback control loop  100  that the sample voltage  14  is compared by amplifier  13  to the reference voltage  15 , so as to form an error correction voltage at the output of amplifier  13 . This is better depicted by FIG. 2. For convenience the sample voltage  14  of FIG. 1 is referred to as Vsam in FIG. 2, and the reference voltage  15  of FIG. 1 is referred to as Vref in FIG. 2. Amplifier  13  can be any type of amplifier circuit and is shown in idealized form. Amplifier  13  is characterized by a voltage gain Ava. Sampler  11  is characterized by a voltage gain Avs, typically less than unity. Voltage reference  12  produces a voltage Vref at the non-inverting input of amplifier  13 . Sampler  11  produces a voltage Vsam at the inverting input of amplifier  13 . The voltages Vref, Vsam, Verrcor, and Vout are taken with respect to node  17 . These inputs are marked “+” and “−,” respectively. The response of amplifier  13  is to produce an output error correction voltage Verrcor: 
           Verrcor=Av *( Vref−Vsam )  Equation 1 
         [0050]    It can be seen from equation 1 that the greater the difference between the reference voltage and the sample voltage, the greater the output error correction voltage generated by the amplifier. FIG. 2 also shows the voltage Vps of the power source and the voltage Vout of the output  17 . By direct substitution: 
           Vout=Vps+Verrcor   Equation 2 
           Vsam=Avs*Vout   Equation 3 
           Vout=Vps+Ava* ( Vref−Avs*Vout )  Equation 4 
         [0051]    Solving for Vout yields: 
           Vout= ( Vps+Ava*Vref )/(1+ Ava*Avs )  Equation 5 
         [0052]    If the amplifier gain is high such that: 
         Inequality 1:  Ava*Vref&gt;&gt;Vps   
         Inequality 2:  Ava*Avs&gt;&gt; 1 
         [0053]    then Vout may be approximated by: 
           Vout   ˜ (1/ Avs )* Vref   Equation 6 
         [0054]    It can be seen that a requirement for the feedback loop to provide Vout as a scaled replica of the Vref is that the amplifier gain be high according to Inequalities 1 and 2. It can also be seen that the reciprocal of the sampler gain Avs is the scale factor between Vout and Vref, according to Equation 6.  
         [0055]    In one preferred embodiment of the present invention the values discussed above are as follows: 
           Ava   ˜ 100,000 
           Avs= 0.033 
           Vref   ˜ 0.033* Vps   
         [0056]    Substituting these values into Inequality 1 gives: 
         (100,000)*(0.033* Vps )&gt;&gt; Vps   
         3,300&gt;&gt;1 
         [0057]    It can be seen that this preferred embodiment meets the requirement for amplifier gain. The intent of the present invention is best served where the product of amplifier gain and sampler gain is at least 10.  
         [0058]    [0058]FIG. 3 shows a combination of two feedback loops  100  and  101 , where the power source  200  of FIGS.  1 - 2  is replaced by a transformer  201  that receives power into a primary winding  210  from an external power source. The entire circuit forms an embodiment of a balanced power conditioner of the present invention. It is composed of an isolation transformer  201  having a primary winding  210  and two identical secondary windings  220  and  230 . Secondary winding  220  has terminals  221  and  222 , so that the voltage at terminal  221  relative to the voltage at terminal  222  is in phase with the voltage at terminal  211  of primary winding  210  relative to the voltage at terminal  212 . Secondary winding  230  has terminals  231  and  232 , so that the voltage at terminal  231  relative to the voltage at terminal  232  is out of phase with the voltage at terminal  211  of primary winding  210  relative to the voltage at terminal  212 .  
         [0059]    A first feedback control loop  100  is tied to secondary  220  and a second loop  101  is tied to secondary  230  to form the power conditioner. A single voltage reference  12  supplies a reference voltage for both feedback loops. More exactly, voltage reference  12  has output  301  which is in phase with the voltage at terminal  221  of secondary winding  220  relative to the voltage at terminal  222 . Voltage reference  12  also has output  302  which is in phase with the voltage at terminal  231  of secondary winding  230  relative to the voltage at terminal  232 . The voltage at output  301  is 180 degrees out of phase with the voltage at output  302 . Feedback control loop  100  provides output  501  of the power conditioner, while loop  101  provides output  503  of the power conditioner. Both loops are grounded to output  502  of the power conditioner, which forms the ground of the output balanced power. The power conditioner of FIG. 3 provides balanced power from either a balanced AC power line or an unbalanced AC power line. The turns ratio of transformer  201  is not specific to the invention and can be determined to provide voltage step up or voltage step down, as well as one-to-one voltage transformation. For example, in one embodiment of the system the turns ratio is one turn of secondary windings  220  and  230  per two turns of primary winding  210 . Additional secondary windings may be added to transformer  201  to provide auxiliary power to the system, such as power for operating the amplifier and reference circuits.  
         [0060]    [0060]FIG. 4 shows an alternate configuration of the present invention, wherein a two wire output is provided. This type of output is normally referred to as unbalanced because the ground of the load network is normally connected to either one of output  501  or output  502 . A transformer  201  having a primary winding  210  and a secondary winding  220  couples the power source main voltage between Input  211  and Input  212  to the servo  100 .  
         [0061]    The use of a transformer in the circuits of FIG. 3 and FIG. 4 provides a means of summing the error correction voltage and power source main voltage.  
         [0062]    [0062]FIG. 5 shows the schematic diagram for the sampler of the preferred embodiment, having an input port  504 , an output port  505 , and a ground port  506 . It can be seen to be a simple voltage divider. The purpose of the sampler is to provide a scaled replica of the output voltage of similar magnitude to that of the voltage reference. The sampler of FIG. 5 is shown to have adjustable gain by variable resister  23 , as a means of adjusting the output voltage of the power conditioner. The voltage gain Avs of the sampler is given below: 
           Avs= ( R   2 + R   3 )/( R   1 + R   2 + R   3 )  Equation 7 
         [0063]    It should be appreciated that a variety of sampling circuit topologies exist that are suitable to providing the scaled sampling function, and that simple voltage dividers may take the form of capacitive networks. Further, an active network could be used in this capacity. In general, the sampler may also provide an arbitrary phase shift or other phase characteristic to the sampled voltage. It is also recognized that the sampler could provide a current output to the amplifier, instead of a voltage. The type of output required from the sampler by the amplifier depends on the amplifier input requirement. If the amplifier is a trans-voltage amplifier then a voltage output is required of the sampler. If the amplifier is a trans-impedence amplifier then a current output is required of the sampler.  
         [0064]    [0064]FIG. 6 depicts an embodiment of the amplifier of the preferred embodiment. It can be seen to be a differential amplifier having an inverting input  507  and a non-inverting input  508 , and a single high current output  509 . In the preferred embodiment an integrated circuit operational amplifier (OpAmp)  52  is followed by a push-pull high current output stage  110 . The integrated amplifier is preferably of the type suitable as a self contained high power audio amplifier capable of delivering around 50 watts of power into an 8 ohm load, such as the National Semiconductor LM3875. The plus supply  512  and the minus supply  513  voltages (voltage rails) of the OpAmp should desireably be at least as large as the maximum anticipated error in the incoming AC sinewave. The preferred embodiment for  117  VAC line voltage applications provides +28 volts and −28 volts to the OpAmp at  512  and  513 , respectively. The preferred embodiment also provides voltage rails of +18 volts and −18 volts to output stage voltage rails  511  and  510 , respectively.  
         [0065]    [0065]FIG. 6 shows an embodiment of the output stage  110  as consisting of four PNP transistors,  24 - 27 , and four NPN transistors,  28 - 31 , wired to form a complementary push-pull amplifier. Emitter resisters  32 - 39  serve to stabilize the bias and promote equitable sharing of bias current over temperature changes. The output stage  110  is base biased by resisters  40  and  45 , and unneeded current is bypassed by transistor  41  and resisters  42 - 44 . Optional to the circut, the bias control circuit comprised of transistor  41  and its associated resisters  42 - 44  form a standard temperature compensated bias network common to the art. Resistor  43  is adjustable to permit trimming of the operating point current in the output transistors. The output of OpAmp  52  is coupled to the output stage  110  through a network consisting of capacitors  46  and  47 , resistors  48  and  49 , and diodes  50  and  51 , as can be seen in FIG. 6. The amplifier of FIG. 6 is suitable for 15 amperes of output current.  
         [0066]    [0066]FIG. 7 shows an alternate topology for the amplifier, suitable for lower current applications in the range of 3-5 amperes. In this embodiment the amplifier  110  consists of only a single branch of transistors  24  and  28 , and their associated bias circuitry. The applications include power supplies for large television receivers and other medium current component electronics.  
         [0067]    [0067]FIG. 8 shows the amplifier as consisting of only OpAmp  52 . The amplifier of FIG. 8 is appropriate when only 1-2 amperes of output current is required. An advantage of this embodiment of the amplifier is to provide a series active power line conditioner that can be integrated into a single function self contained apparatus such as a compact disc player.  
         [0068]    In each case the voltage rails for the amplifier are small compared to the magnitude of output AC voltage of the power conditioner, this made possible by the fact that only the relatively small error correction voltage is generated by the amplifier. This stands in contradistinction to the power regenerative schemes of the prior art, wherein the amplifier must generate the entire AC output voltage of the power conditioner, as in the embodiments of FIG. 9. For this reason there is a considerable improvement in efficiency compared with power regenerative systems. The efficiency of the preferred embodiment of the present invention is about 75%.  
         [0069]    The voltage reference in the embodiment is derived from the pre-conditioned AC power by sampling and filtering. This differs from the prior art in that no oscillator and phase locked loop is required to ensure a reference voltage that is phase coherent to the incoming AC power. FIG. 10 shows an embodiment in which the voltage reference is a sampling comparator  62 , followed by a filter  63 . FIG. 11 shows a block diagram of a system for generating a voltage reference of the preferred embodiment. The reference  12  is composed of sampling comparator  62 , active low pass filters  64  and  65 , passive low pass filter  66 , and phase splitter  67 . The phase splitter is used when balanced power output is generated from the power conditioner. The active low pass filters of the preferred embodiment are configured as 8th order low pass filters. Other types of low pass filters may be used, such as 2nd order or higher order low pass filters. Alteratively, bandpass filters may be used instead of low pass filters.  
         [0070]    The sampling comparator receives an AC voltage from the incoming AC power source. One possible configuration for AC sampling is shown in FIG. 12, where the AC voltage is sampled from a secondary winding  252  of the isolation transformer  250  of the power conditioner. Shown as secondary  252  in the figure, the AC voltage could be sampled from any convenient secondary tap of the transformer.  
         [0071]    Referring to FIG. 13, the sampled AC voltage at the Input  74  of the sampling comparator  73  is AC coupled through capacitor  79  to a voltage divider consisting of resistors  80  and  82 . An adjustable phase lag circuit comprised of resistor  81  and capacitor  83  provide phase optimization of the feedback control loop. The signal is then coupled to the non-inverting input of a high gain rail-to-rail OpAmp  90 , which is an OpAmp capable of generating an output voltage anywhere within the range of its power supply voltages. The signal emerges from OpAmp  90  as a squarewave of amplitude determined by the rail voltages presented to OpAmp  90 . Over 95% of the random noise power content on the incoming AC power sampled by this circuit is stripped off by OpAmp  90 , and is replaced by the well defined harmonic structure of a squarewave. This approach is an improvement to prior art filtering techniques which simply low pass filter the AC signal using linear filtering technology.  
         [0072]    The positive rail voltage for OpAmp  90  is provided by OpAmp  89 , which is configured to operate as a voltage buffer. A +15V supply voltage is divided down to about 2.7V by resisters  88  and  87 . Capacitor  86  provides noise filtration of the +2.7V input to OpAmp  89 . In a similar manner, the negative voltage rail to OpAmp  90  is provided by OpAmp  91 , resistors  92  and  84 , and capacitor  85 . The output from OpAmp  90  is AC coupled to the output port  78  of the comparator through capacitor  93 .  
         [0073]    [0073]FIG. 14 shows an alternate configuration of the sampling comparator. In this configuration the voltage rails for OpAmp  90  are allowed to track the long term average value of the incoming peak AC voltage. In this case long term average means an average taken over ten to several hundred cycles. This causes the reference voltage and the output voltage of the power conditioner to track the average incoming AC voltage. Short term (less than one period) transient errors in supply voltage do not appear at the operating rails of OpAmp  90 , due to the filtering capacitors  86  and  85 . The time constant of capacitor  86  with resistors  87  and  88  is chosen to be long enough to prevent transient activity and noise on the incoming power from passing through OpAmp  89 . The time constant is also chosen short enough to allow the output of OpAmp  89  to track the long term average voltage of the incoming power. The time constant is preferably in the range of 1-10 seconds. OpAmp  91  and its components operate in a similar manner. As a consequence, the operating voltage rails for the amplifier are allowed to be reduced to the minimum values consistent with the expected maximum short term voltage error of the incoming AC power. No further voltage headroom need be allowed for average voltage line regulation. This kind of regulation may be referred to as transient line regulation, as only transient errors in supply voltage are corrected. The benefit of this practice is significantly improved efficiency, on the order of 85%, due to the lower magnitude voltage rails on the amplifier.  
         [0074]    [0074]FIG. 15 shows the configuration of a standard continuous time filter integrated circuit to form the 8th order butterworth low pass filter of the present invention. This is merely an example of the type of filter that could be used. The IC of the preferred embodiment is a MAX274 produced by Maxim Integrated Products. This IC is divided into four identical sections, each one of which is configured by the associated external resisters to form independent sections of the butterworth filter. Section  1  is programmed by resisters  601 - 608 . Section  2  is programmed by resisters  609 - 616 . Section  3  is programmed by resisters  617 - 624 . Section  4  is programmed by resisters  625 - 632 . Capacitors  635 - 638  AC couple the various sections together so as to prevent any DC offsets generated by the filter sections from accumulating. The filter IC is biased from +5V and −5V supplies through RC low pass filter networks comprised of resister  634  and capacitor  640 , and resistor  633  and capacitor  639 , respectively.  
         [0075]    Numerous other filter structures common to the art are possible substitutions for the filter described in FIG. 15, including passive filters. Any filter type that provides at least 20 dB of suppression of the third and higher harmonics relative to the fundamental frequency of the incoming AC is a useful substitution for the filter described in FIG. 15.  
         [0076]    [0076]FIG. 16 illustrates an embodiment of the passive filter of the present invention. It is a three section filter composed of RC sections. A first section is comprised of resister  651  and capacitor  654 . A second section is comprised of resister  652  and capacitor  655 . A third section is comprised of resister  653  and capacitor  656 . Each section forms a 6 dB per octave low pass filter. The pole frequency of each low pass section is chosen to be between 10 and 1000 times the frequency of the incoming AC power. The main purpose of this filter is to suppress any noise generated by the active filters that precede it.  
         [0077]    [0077]FIG. 17 depicts an embodiment of the phase splitter of the preferred embodiment of the present invention. The phase splitter is composed of a pair of OpAmps. OpAmp  662  is wired with its output directly connected to its inverting input. The non-inverting input serves as the input of the resultant non-inverting unity gain buffer. OpAmp  663  is wired with a first resister  664  of value R connected between its output and its inverting input. The non-inverting input is grounded. A second resister  665 , also of value R, is connected between the inverting input of OpAmp  663  and the non-inverting input of OpAmp  662 . OpAmp  663  operates as an inverting unity gain buffer. The entire circuit operates as a phase splitter, where the non-inverting input of OpAmp  662  is the input of the phase splitter and the output of OpAmp  660  and OpAmp  661  serve as the non-inverting output and the inverting output, respectively.  
         [0078]    [0078]FIG. 18 shows a power conditioner for unbalanced output power. Transformer  670  provides secondary voltage from secondary winding  671  to operate in conjunction with feedback control loop  100  to provide unbalanced output voltage at output  501  with respect to the output voltage at output  502 . In addition, transformer  670  provides the power for the amplifier of feedback control loop  100  from secondary winding  672 . Diodes  675  and  676 , along with capacitors  677  and  678 , perform the rectification of voltage from secondary winding  672 , whereby the voltage for the amplifier voltage rails is provided. Auxiliary power for the voltage reference circuitry is provided by an auxiliary power supply  679  and by secondary winding  673 .  
         [0079]    [0079]FIG. 19 depicts a balanced power preferred embodiment of the present invention, in which secondary winding  681  provides power to feedback control loop  100  to produce an output voltage at output  501 , while secondary winding  684  provides power to feedback control loop  101  to produce an output voltage at output  503 , 180 degrees out of phase from the voltage at output  501 . The feedback control loops are tied together at output  502 , which forms the ground terminal of the balanced power output. Amplifiers  694  and  695  are powered by a balanced power supply consisting of center tapped secondary winding  682  and  683 , a full wave rectifier formed by diodes  687 - 690 , and filter capacitors  691  and  692 . The center tap is tied to output ground  502 .  
         [0080]    An advantage of this embodiment of the system is to provide balanced power from a balanced or unbalanced power line.  
         [0081]    Several improvements and enhancements to the embodiments described above will result in improved stability of the feedback control loop and improved tolerance to a wide variety of load impedances and corresponding power factors. Some of these are described below.  
         [0082]    [0082]FIG. 20 shows the output stage  110  of the amplifier as consisting of a plurality of series R-C networks tied across the emitter and collector terminals of the output transistors. Resistor  701  is in series with capacitor  709  to form a series network that is connected across output transistor  24  from emitter to collector. In a similar fashion, a series R-C network appears across each of the output transistor. This technique is useful in stabilizing the servo without compromising operating bandwidth. Typical operating bandwidth is about 100 KHz.  
         [0083]    [0083]FIG. 21 shows a method of connecting the sampler  800  and the voltage reference  801  to the amplifier  720  of the feedback control loop  120 , in which the sampler  800  feeds terminal  724  of coaxial transmission line  723 , the corresponding shield connection  726  is made to node  502 . Terminal  722  of transmission line  723  feeds the inverting input  721  of amplifier  720 . The voltage reference  801  feeds terminal  731  of coaxial transmission line  729 , the corresponding shield connection  733  is made to node  502 . Terminal  730  of transmission line  729  feeds the non-inverting input  727  of amplifier  720 . The shield terminal  732  of transmission line  729  is connected to the shield terminal  725  of transmission line  723 . Both terminals  725  and  732  are connected to amplifier input  727  through capacitor  728 . This arrangement is found to preserve the high frequency stability of the feedback control loop and to reject noise that may otherwise enter the system through finite line lengths connecting the sampler  800  and the voltage reference  801  to the amplifier  720 .  
         [0084]    [0084]FIG. 22 shows an embodiment of a series R-C network formed by capacitor  750  and resistor  751 ; the network thus formed is connected across the secondary winding  671  of isolation transformer  670 . The series R-C network is useful in stabilizing the feedback control loop  101  over a broad frequency range.  
         [0085]    [0085]FIG. 23 illustrates an embodiment of a series R-C network formed by resistor  681  and capacitor  680 , the series network connected between the output terminal  501  of the power conditioner and the inverting input  14  of the amplifier  13  so as to promote stability of the feedback control loop  100 .  
         [0086]    [0086]FIG. 24 shows an embodiment of a series R-C network formed by resistor  682  and capacitor  683 , the series network connected across the output of the power conditioner between output terminals  501  and  502  so as to stabilize the feedback control loop to a variety of load impedances.  
         [0087]    There are no doubt several alternative embodiments that result in a power conditioner operating in accordance with the descriptions presented herein. Those skilled in the art will appreciate that such alternatives do not obscure the intent of the present invention, nor do they transcend the spirit of the invention hereby disclosed.