Abstract:
A new spread spectrum phase modulation (SSPM) technique is applicable to both data and clock signals. The SSPM technique is more suitable to board level designs than the direct-sequence spread spectrum (DSSS) technique. In addition, SSPM may be combined with controlled edge rate signaling to outperform DSSS.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application claims the benefit of prior filed copending provisional application Ser. No. 60/071,805, titled “Suppression of Electromagnetic Interference in Parallel Data Channels through Spread Spectrum Phase Modulation,” filed on Jan. 20, 1998 by inventors Yongsam Moon, Deog-Kyoon Jeong, and Gyudong Kim. 
    
    
     BACKGROUND OF THE INVENTION 
     1. Technical Field 
     The present invention relates generally to electronic circuitry for parallel clock and data transmission. More particularly, the present invention relates to reducing electromagnetic interference (EMI) during such transmission. 
     2. Description of Related Art 
     As electronic and computer technology continues to evolve, communication of data among different devices, either situated nearby or at a distance, becomes increasingly important. It is also increasingly desirable to provide such data communications at very high speeds, especially in view of the large amount of data required for data communications in intensive data consuming systems using graphical or video information, multiple input-output channels, local area networks, and the like. Hence, it is now more desirable than ever to provide for high speed data communications among different chips on a circuit board, different circuit boards in a system, and different systems with each other. 
     A problem of increasing significance for such data communications is substantial electromagnetic interference (EMI) radiation, often exceeding acceptable levels. As the number of data lines and the rate of data driving and transmission increases, the EMI emitted increases correspondingly. 
     An early prior art method of reducing EMI radiation involves physical shielding. Physical shielding may reduce EMI radiation, but physical shielding may be cumbersome and costly, and may not be effective enough to sufficiently reduce EMI radiation depending on the frequencies involved. 
     Electromagnetic interference may have an adverse influence on the operations of electronic equipment. Thus, there are strict regulations on electromagnetic emission covering both industrial and consumer electronic equipment. Recently, there is increasing pressure to reduce EMI from such equipment. 
     An on-board parallel clock and data channel as shown by the example in FIG. 1 is a primary source of EMI for some systems. In the following analysis, we assume a dual edge clocking scheme for simplicity and since it is more favorable to the EMI problem. In the far-field, each metal wire may be considered as a single point, and the EMI power radiated by the wire trace is calculated as P(f)∝I 2 (f)·f 2 , where f is the signal frequency and I(f) is the current through the wire. For example, assuming that 8 bit data wires carry an identical alternating 01 sequence with a clock of 62.5 megahertz (MHz) with rising and falling times of 1 nanosecond (ns), an EMI peak occurs at 812.5 MHz as shown in FIG.  2 ( c ). Note that only the current waveform shown in FIG.  2 ( b ) is related with EMI rather than the voltage waveform shown in FIG.  2 ( a ). 
     In order to reduce the peak EMI, either the power spectrum of EMI must be evenly spread over a wide frequency range or high frequency components of the current must be reduced. 
     One of the conventional techniques is direct-sequence spread spectrum (DSSS), where each data is exor&#39;ed with a pseudo-random sequence and then exor&#39;ed with the same sequence to recover data in the receiver. This spreads the data in frequency prior to transmission and “despreads” it at the receiver, as shown by the example illustrated in FIG.  3 . 
     However, the DSSS technique has a substantial disadvantages and problems. One disadvantage is that the DSSS technique can be applied to data signals, but not to a clock signal. This is because the clock signal must be glitch and jitter free. In the example shown in FIG. 3, the EMI reduction is merely to negative 19.1 dB (decibels) at 812.5 MHz, and the remaining peak arises primarily from the unspread clock line. [1 dB=10 log 10  (P 2 /P 1 ), where P 1  and P 2  represent the power of two signals.] One of the problems is that the DSSS technique requires pseudo-random (PN) code generators in both transmitter and receiver for scrambling/descrambling and synchronization between transmitter and receiver. 
     SUMMARY OF THE INVENTION 
     The above described problems and disadvantages are overcome by the present invention. The present invention relates to a new spread spectrum phase modulation (SSPM) technique that is applicable to both data and clock signals. The SSPM technique is more suitable to board level designs than the direct-sequence spread spectrum (DSSS) technique. In addition, SSPM may be combined with controlled edge rate signaling to outperform DSSS. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a schematic diagram showing a typical configuration including a transmitter, a receiver, and a channel comprising a clock line and 8 data lines. 
     FIG.  2 ( a ) is a graph illustrating a voltage waveform output by a pad of a transmitter to a wire of a channel. 
     FIG.  2 ( b ) is a graph illustrating a current waveform output by a pad of a transmitter to a wire of a channel. 
     FIG.  2 ( c ) is a graph illustrating a power spectrum due to the current waveform of FIG.  2 ( b ). 
     FIG.  3 ( a ) is a schematic diagram showing a direct-sequence spread spectrum communication system, including pseudo-random code generators within a transmitter and a receiver. 
     FIG.  3 ( b ) is a graph illustrating the spreading of a data signal and the non-spreading of a clock signal by way of the direct-sequence spread spectrum technique. 
     FIG.  3 ( c ) is a graph illustrating the reduction of the peak values in the power spectrum when the direct-sequence spread spectrum technique is applied. 
     FIG.  4 ( a ) is a graph illustrating phase modulation of a signal in accordance with a preferred embodiment of the present invention. 
     FIG.  4 ( b ) is a graph illustrating the phase of the signal dithered by a pseudo-random code in accordance with a preferred embodiment of the present invention. 
     FIG.  5 ( a ) is a schematic diagram illustrating a spread spectrum phase modulation communication system in accordance with a preferred embodiment of the present invention. 
     FIG.  5 ( b ) is a graph illustrating the improved reduction of the peak values in the power spectrum when the spread spectrum phase modulation technique is applied in accordance with a preferred embodiment of the present invention. 
     FIG.  6 ( a ) is a graph illustrating an output voltage waveform having an increased transition time in accordance with a preferred embodiment of the present invention. 
     FIG.  6 ( b ) is a graph illustrating an output current waveform having an increased transition time in accordance with a preferred embodiment of the present invention. 
     FIG.  6 ( c ) is a graph illustrating the further improved reduction of the peak values in the power spectrum when the transition time is increased and the spread spectrum phase modulation technique is applied in accordance with a preferred embodiment of the present invention. 
     FIG. 7 is a schematic diagram showing SSPM transmitter circuitry in accordance with a preferred embodiment of the present invention. 
     FIG.  8 ( a ) is a schematic diagram showing circuitry for a T/2 Phase Detector in accordance with a preferred embodiment of the present invention. 
     FIG.  8 ( b ) is a graph illustrating clock and phase detection signals in accordance with a preferred embodiment of the present invention. 
     FIG.  8 ( c ) is a graph of phase difference vs. control voltage variation in accordance with a preferred embodiment of the present invention. 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Spread Spectrum Phase Modulation and EMI Reduction 
     FIG.  4 ( a ) shows a signal waveform under phase modulation. The original and unmodulated signal  402  is shown in the top line of FIG.  4 ( a ). The phase modulated, or dithered, signal  404  and its phase  406  are shown in the second and third lines of FIG.  4 ( a ). 
     As shown, the phase  406  varies continuously between 0 degrees (EARLY state) and negative 180 degrees (LATE state). To prevent excessive phase change between the two successive phase values (EARLY state and LATE state), a SLOW state (EARLY to LATE transition) and a FAST state (LATE to EARLY transition) are inserted between transitions to and from EARLY and LATE states. In accordance with a preferred embodiment of the present invention, the SLOW and FAST states occupy at least 16 cycles, and the phase change between two successive cycles is limited to 12 degrees. Of course, within the scope of the present invention, the number of cycles occupied and the phase change between two successive cycles may vary from the particular numbers above. 
     FIG.  4 ( b ) is a graph illustrating the phase  408  of the signal dithered by a pseudo-random code (PN sequence)  410  in accordance with a preferred embodiment of the present invention. For purposes of illustration, the pseudo-random sequence  410  shown starts with the sequence 011010. Techniques for generating such pseudo-random sequences are known to those of ordinary skill in the pertinent art. 
     When the phase modulation is controlled by a PN sequence  410  such as shown in FIG.  4 ( b ), the resultant power spectrum will be spread like the power spectrum in FIG.  5 ( b ). The power spectrum in FIG.  5 ( b ) has peaks with a maximum power of negative 14.6 dB 1 GHz. In comparison, the power spectrum in FIG.  2 ( b ) has peaks with a maximum power of 0 dB. Thus, applying spread spectrum phase modulation in this way to the signal results in a magnitude 14.6 dB reduction in peak EMI. 
     Although the 14.6 dB reduction from this implementation of SSPM is substantial, it is less than the 19.1 dB reduction from the implementation of DSSS shown in FIG.  3 ( c ). Nevertheless, this implementation of SSPM is advantageous over DSSS because, unlike DSSS, SSPM does not require a pseudo-random code generator in the receiver and so requires simpler circuitry in comparison to the circuitry for DSSS shown in FIG.  3 ( a ). 
     A SSPM transmitter circuit  502  for parallel transmission of a clock signal and multiple data signals and for phase modulation of those clock and data signals is shown in FIG.  5 ( a ). The circuit  502  includes: a clock signal source  504  for generating the clock signal (CLK); a plurality of data signal sources  506  for generating the multiple data signals (D 0 , D 1 , D 2 , . . . , D 7 ); a control voltage source  508  for generating a control voltage (Vctrl 1 ); a first voltage-controlled delay line  510  coupled to the clock signal source  504  to receive the clock signal and coupled to the control voltage source  508  to receive the control voltage, the first voltage-controlled delay line delaying the clock signal according to the control voltage; and a plurality of voltage-controlled delay lines  512  coupled to the plurality of data signal sources  506  to receive the multiple data signals and coupled to the control voltage source  508  to receive the control voltage, the plurality of voltage-controlled delay lines delaying the multiple data signals according to the control voltage. The output of the circuit  502  is also shown in FIG.  5 ( a ). The first voltage-controlled delay line  510  outputs a dithered clock (dithered CLK) signal  514 . The plurality of voltage-controlled delay lines  512  output dithered data signals  516 . 
     Thus, the spread spectrum phase modulation (SSPM) technique can be applied to both clock and data without skew errors between data and clock as shown in FIG.  5 ( a ). The absence of skew errors is achieved by phase-modulating the clock and data through voltage-controlled delay lines (VCDLs  510  and  512 ) of which delays are controlled by the same control voltage. It is desirable that the phase difference between maximum and minimum delays applied by the VCDLs should be 180 degrees. This is because as the phase difference between maximum and minimum delays gets away from 180 degrees, the EMI reduction gets smaller according to our simulations. 
     Effect of Increased Transition Time (ITT) of Data Outputs 
     In order to reduce the high frequency component of the current, increasing the transition time (t s ) is desirable. However, the slow edge rate cannot be applied to a clock signal, so EMI reduction on a clock signal is not expected. 
     Since the negative 19.1 dB peak at 812.5 MHz in the case of DSSS is due primarily to the clock signal, no further peak reduction would occur by increasing the transition time (t s ) in the case of DSSS. In contrast, since the negative 14.6 dB peak at 1 GHz in the case of SSPM is not due primarily to the clock signal, that peak will be substantially further reduced occur by increasing the transition time (t s ) in the case of SSPM. 
     FIG.  6 ( a ) is a graph illustrating an output voltage waveform having an increased transition time (t s ) in accordance with a preferred embodiment of the present invention. The increased transition time (t s ) is more distinctly shown in FIG.  6 ( b ) which shows the corresponding output current waveform. The transition time (t s ) for the waveforms shown in FIGS.  6 ( a ) and  6 ( b ) is 5 nanoseconds (ns). In comparison, the transition time (t s ) for the waveforms shown in FIGS.  2 ( a ) and  2 ( b ) is 1 nanosecond (ns). 
     FIG.  6 ( c ) is a graph illustrating the further improved reduction of the peak values in the power spectrum when the transition time (t s ) is increased to 5 ns, and the spread spectrum phase modulation technique is applied in accordance with a preferred embodiment of the present invention. As can be seen from FIG.  6 ( c ), the peak at 1 GHz is further reduced to negative 31.3 dB. 
     FIG. 7 is a schematic diagram showing SSPM transmitter circuitry  700  in accordance with a preferred embodiment of the present invention. The transmitter circuitry  700  includes the phase selection circuit (PSC)  508  and a delay lock loop (DLL)  702 . Both the PSC  508  and the DLL  702  supply control voltages to a voltage-controlled delay line (VCDL)  510 . The same or similar circuitry would be used to supply control voltages to the other voltage-controlled delay lines  512 . 
     The transmitted signal (the CLK signal in the instance shown in FIG. 7) is modulated by the VCDL  510 . The delay applied by the VCDL  510  is controlled by two control voltages: Vctrl 1  and Vctrl 2 . 
     The generation of Vctrl 1  by the PSC  508  is controlled by a switching algorithm, and Vctrl 1  is used for interpolating the delay applied by the VCDL  510 . For example, the VCDL  510  generates a minimum delay (0) when Vctrl 1  is switched to V 15 . As another example, the VCDL  510  generates a maximum delay (T/2) when Vctrl 1  is switched to V 0 . According to a preferred embodiment of the present invention, Vctrl 1  is continuously switched from V 15  to V 14 , V 13 , V 12 , and so on to V 0 , then to V 1 , V 2 , V 3 , and so on to V 15 , etc. 
     The DLL  702  generates Vctrl 2  corresponding to a half period (T/2) delay difference. The DLL  702  includes a T/2Phase Detector  704  with CLK 0  and CLK 1  input signals, and UP and DOWN output signals. The DLL  702  adjust s Vctrl 2  until a rising edge of the CLK 0  signal and the falling edge of the CLK 1  signal are aligned. 
     As Vctrl 1  is continuously switched between V 15  and V 0  according to the switching algorithm using a pseudo-random sequence  410 , the delay applied by the VCDL  510  varies between 0 and T/2. Furthermore, because a low-pass filter  706  is used in the generation of Vctrl 1 , the phase and delay vary smoothly. 
     FIG.  8 ( a ) is a schematic diagram showing circuitry for a T/2 Phase Detector  704  in accordance with a preferred embodiment of the present invention. The T/2 Phase Detector  704  comprises a dynamic phase detector that has two input signals CLK 0  and CLK 1  and two output signals UP and DOWN. 
     For generating the UP signal output, the CLK 1  signal is input to a first inverter  802  and to gates of a first PMOS transistor  804  and a first NMOS transistor  806 . The source of the first PMOS transistor  804  is coupled to a supply voltage, and the drain of the first PMOS transistor  804  is coupled to the source of a second PMOS transistor  808 . The source of the first NMOS transistor  806  is coupled to the drain of the second PMOS transistor  808 , and the drain of the first NMOS transistor  806  is coupled to an electrical ground. The CLK 0  signal is input to a second inverter  810 . 
     In addition, the output of the first inverter  802  is coupled to a gate of a third PMOS transistor  812 . The output of the second inverter  810  and the gate of the second PMOS transistor  808  are coupled to a gate of a second NMOS transistor  814 . The node between the drain of the second PMOS transistor  808  and the source of the first NMOS transistor  806  is coupled to the gate of a third NMOS transistor  816 . 
     Furthermore, the source of the third PMOS transistor  812  is coupled to a supply voltage, and the drain of the third PMOS transistor is coupled to an input of a third inverter  818 . The source of the second NMOS transistor  814  is also coupled to the input of the third inverter  818 , and the drain of the second NMOS transistor  814  is coupled to the source of the third NMOS transistor  816 . The drain of the third NMOS transistor  816  is coupled to an electrical ground. Finally, the output of the third inverter  818  comprises the UP output signal. 
     For generating the DOWN signal output, the circuitry is the same as that for generating the UP signal, except that the CLK 0  and CLK 1  input signals are reversed as shown in the bottom half of FIG.  8 ( a ). 
     The circuitry shown in FIG.  8 ( a ) comprises a dynamic phase detector with fewer transistors and higher precision than prior dynamic phase detectors. Owing to the high precision of its dynamic logic operation, the T/2 Phase Detector  704  can operate without any phase offset. 
     FIG.  8 ( b ) is a graph illustrating clock and phase detection signals in accordance with a preferred embodiment of the present invention. As shown in FIG.  8 ( b ), the widths of UP and DOWN pulses are proportional to the phase difference of the inputs CLK 0  and CLK 1 . Further, there are no pulses in lock state. 
     FIG.  8 ( c ) is a graph of phase difference vs. control voltage variation in accordance with a preferred embodiment of the present invention.