Abstract:
This disclosure relates to a programmable wideband, LC Tuned, Voltage Controlled Oscillator with continuous center frequency select, and independent configuration of amplitude and tuning gain. The programmability can be via on chip non-volatile memory, or through data shifted into the part and stored via a data bus.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to voltage controlled oscillators (VCOs), particularly wideband VCOs. 
     2. State of the Art 
       FIG. 1  shows typical complementary LC tuned VCO  100  with a single tuning port configured for wideband operation. A complementary cross coupled CMOS inverter is formed by PMOS transistors M 1 , M 2  and NMOS transistors M 3 , M 4 . The PMOS transistors are cross coupled such that a gate electrode of each is coupled to a drain electrode of the opposite PMOS transistor. Similarly, the NMOS transistors are cross coupled such that a gate electrode of each is connected to a drain electrode of the opposite NMOS transistor. The PMOS transistor M 1  and the NMOS transistor M 3  are coupled drain to drain, and the PMOS transistor M 2  and the NMOS transistor M 4  are coupled drain to drain. Source electrodes of the PMOS transistors are connected to a supply voltage VDD. Source electrodes of the NMOS transistors are connected through a tail current source I to ground. Complementary output signals are formed between the PMOS transistor M 2  and the NMOS transistor M 4 , on the one hand (output terminal C), and between the PMOS transistor M 1  and the NMOS transistor M 3 , on the other hand (output terminal CZ). Coupled between the output terminals are frequency controlling reactive elements including an inductor L, a capacitor C and varactors V and V z . A tuning input signal (TUNING) is connected to control terminals of the varactors at node X. 
     Wide band tuning implies high tuning gain which is undesirable for noise considerations, as may be appreciated from the following example. For a VCO to cover the range from 800 MHz to 1700 MHz, at 3V operation, the tuning gain will be 300 MHz/V. In this instance, 1 mv of noise on the tuning node will translate to 300 kHz of frequency deviation (phase noise). 
       FIG. 2  shows a typical complementary LC tuned VCO with a digital, “coarse” tuning port which divides the band into a series of sub-bands, and an analog, “fine” tuning port which functions as the tuning port for a PLL. In this case, since the PLL only operates in a sub-band, the tuning gain is reduced to the sub-band width divided by the supply. 
       FIG. 3  shows the frequency breakdown of the prior art VCO of  FIG. 2  for the case of eight sub-bands. Note that: 
     1. Each coarse selected sub-band must overlap, with fixed varactor. Since all frequencies of the total band must be achievable, there can be no gaps from one selected band to the next, implying each sub-band must have overlap. 
     2. The amount of overlap depends on the number of sub-bands and on process/temperature/voltage variation. If a selected frequency is near the top or bottom of any sub-band, the sub-band overlap must be sufficient such that the tuning voltage can maintain the desired frequency. 
     3. Operation in sub-band overlap typically causes charge pump to operate outside optimum output tuning voltage for lowest spurs (i.e., near either rail) due to finite output resistance of the devices. 
     4. Each coarse tuned sub-band progressively compresses as frequency decreases, with fixed varactor.  FIG. 3  shows the band width for each sub-band using a fixed varactor. The right hand scale shows the tuning gain plotted at each sub-band assuming 3V operation. Note that the gain varies by almost an order of magnitude. 
     5. Sub-band compression forces higher frequency sub-bands to have higher tuning gain, since a minimum gain must be used when designing a PLL loop. As a consequence, there will be only one sub-band with optimum gain. 
     6. Tuning gain is proportionally dependent on center frequency. That is, as the desired frequency increases, the tuning gain increases at approximately the same rate, as shown in  FIG. 3 . 
     Although not illustrated in  FIG. 3 , the VCO of  FIG. 2  experiences amplitude loss across the range. VCO amplitude decreases at lower frequency sub-bands due to increased capacitive loading of the fixed drive amplifier. The amplitude is inversely proportional to load, and proportional to frequency. Automatic amplitude control (AAC) can be used at the cost of increased die area and increased noise, especially within the loop bandwidth of the control circuit. An alternative to AAC is to program the current source when programming frequency. In both the AAC case and the programming case, the use of a current source contributes significant noise to the VCO through channel noise of the source, and the 2x fundamental located at the common node. In order to keep channel noise low, the current source transistors are typically very large, adding to die area. 
     SUMMARY OF THE INVENTION 
     The present invention, generally speaking, provides for a programmable, wideband VCO that has all major variables independent of each other such that it can be configured for any frequency/condition at optimum performance. A number of coarse tuning bits is provided sufficient to achieve continuous center frequency selection, independent of other programming parameters. Programmable varactors are used to achieve tuning gain independent of center frequency. This measure allows tuning gain to be kept at the optimum value regardless of selected center frequency. Programmable transconductance (gm) stages are used to achieve amplitude independent of VDD and center frequency over the total band. This measure allows tuning gm to be kept at the optimum value regardless of selected center frequency. The current source characteristic of the prior art is eliminated, removing two major sources of VCO noise and reducing die area. An integrated active supply filter reduces power supply induced noise. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWING 
       The present invention may be further understood from the following description in conjunction with the appended drawing. In the drawing: 
         FIG. 1  is a circuit diagram of one prior art VCO. 
         FIG. 2  is a circuit diagram of another prior art VCO, wherein the outlined portion (section F) corresponds to section F of  FIG. 5 . 
         FIG. 3  is graph illustrating characteristics of the VCO of  FIG. 2 . 
         FIG. 4  is a graph illustrating characteristics of the VCO of  FIG. 5 . 
         FIG. 5  is a circuit diagram of a VCO in accordance with one embodiment of the invention. 
         FIG. 6  is a circuit diagram of the VCO tuning stage of  FIG. 5 . 
         FIG. 7  is a circuit diagram of the VCO transconductance stage of  FIG. 5 . 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       FIG. 5  shows a block level schematic of the programmable, wideband VCO. Each stage independently controls the frequency, tuning gain, and Gm of the VCO without significantly affecting the other. 
     More particularly, a coarse frequency select stage F, a tuning gain control stage VC, and a variable transconductance output stage G (which includes circuitry like that of the oscillator  100  of  FIG. 1 ) are all connected to the output terminals C, CZ. Each of the stages has an associated control bus (GM_SELECT BUS, FINE_FREQ_SELECT_BUS and VCO_GAIN_SELECT_BUS, respectively.) The voltage control stage VC has a TUNING_VOLTAGE input signal. 
     The frequency select stage F consists of binary weighted capacitors that are switched into the tank through NMOS switches. Coarse frequency control can be achieved in conventional fashion as depicted in  FIG. 2 . As shown therein, a first series of binary weighted capacitors Csbin 1  is formed with one plate of each capacitor being connected to one side of the oscillator configuration  100 . The other plate of the capacitors is connected through a controllable transistor switch (T 1   1 , T 2   1 , etc.) to ground. A second, complementary, series of binary weighted capacitors Csbin 2  is connected to the other side of the oscillator configuration. Capacitors of the same weight on opposite sides of the oscillator configuration are paired together, and their respective transistor switches are commonly controlled. Hence, a 1X_SELECT input signal, when asserted, couples capacitors of weight  1  into the circuit on each side of the oscillator configuration such that the frequency of oscillation is reduced. A 2X_SELECT input signal, when asserted, couples capacitors of weight  2  into the circuit on each side of the oscillator configuration such that the frequency of oscillation is further reduced, etc. 
     The voltage control stage consists of binary weighted varactors that are switched either onto the tuning line or to the cmin or cmax state, as depicted in  FIG. 6 . The cmin/cmax state of unused varactors is determined by the state of the common “unused_varactor_state” bus, which is ground for cmin, and VDD for cmax. The number of cells selected is determined by the gain desired for any given frequency. 
     More particularly, a series of binary weighted varactors VS 1  is formed with one plate of each varactor being connected to one side (C) of the oscillator configuration. The other plate of the respective varactors is connected to respective control voltage nodes N 1 , N 1 , etc. A complementary series of binary weighted varactors VS 2  is connected to the other side (CZ) of the oscillator configuration. Capacitors of the same weight on opposite sides of the oscillator configuration are paired together, and are commonly controlled. Hence, the state of each varactor pair is determined by a voltage applied to the control voltage node. That voltage may be a supply voltage VDD, a reference voltage VSS, or an intermediate control voltage (VOLTAGE_CONTROL) applied by the user. A circuit CTL 1  that determines a voltage applied to the control voltage node N 1  will be described. 
     The control voltage node N 1  is connected to VDD through a pair of PMOS transistors M 1 , M 2 , to VSS through a pair of NMOS transistors M 3 , M 4 , and to a voltage control input signal through a pass gate P. An enable signal ENABLE — 1X is applied in its true form to one side of the pass gate and to the PMOS transistor M 2 . The enable signal is inverted by an inverter INV and is applied in its inverted form to the other side of the pass gate P and to the NMOS transistor M 3 . When the enable signal is asserted, the pass gate is opened, and the VOLTAGE_CONTROL signal is applied to the control voltage node N 1 . At the same time, the PMOS transistor M 2  and the NMOS transistor M 3  are rendered non-conducting. 
     An UNUSED_VARACTOR_STATE signal is applied to the PMOS transistor M 1  and to the NMOS transistor M 4 . Depending on the value of this signal, one of these two transistors is rendered conducting and the other non-conducting. As a result, when the UNUSED_VARACTOR_STATE signal is low, the voltage VDD is applied to the source of the PMOS transistor M 2  while the source of the NMOS transistor M 3  remains floating. When the UNUSED_VARACTOR_STATE signal is high, the voltage VSS is applied to the source of the NMOS transistor M 3  while the drain of the NMOS transistor M 2  remains floating. When the enable signal is deasserted, the voltage determined by the UNUSED_VARACTOR_STATE signal is applied to the control voltage node. 
     Note that UNUSED_VARACTOR_STATE signal is connected in common to all of the varactor pairs. Similarly, the VOLTAGE_CONTROL signal is connected in common to all of the varactor pairs. Hence, if a varactor pair is enabled, it will be controlled by the VOLTAGE_CONTROL signal. If a varactor pair is not enabled, it, along with any and all other varactor pairs not enabled, will be set to either a minimum capacitance state or a maximum capacitance state depending on the UNUSED_VARACTOR_STATE signal. 
     The voltage control stage may be programmed to achieve a desired tuning voltage gain independent of the chosen frequency of operation (i.e., independent of the address number of the fine frequency adjust stage). To illustrate, the frequency of operation is given by the following equation:
 
 F+dF= 1/(2π( L *( C+dC )) 1/2 )  (2)
 
     If it is desired to keep dF constant for any frequency (F), then setting dF equal to a constant and solving for dC will provide an equation modeling the desired behavior of the tuning voltage gain stage. 
     Substituting ½π( LC ) 2  for F to give only two variables:
 
 F= 1/(2π( L*C ) 1/2 )  (3)
 
 F+dF= 1/(2π( L *( C+dC )2π)  (4)
 
( F+dF ) 2 =1/4π 2   *L* ( C+dC )  (5)
 
 C+dC= ¼π 2   *L* ( F+dF ) 2   (6)
 
     Substituting (3) into (6):
 
 C+dC= 1/(4π 2   *L* (½π( L*C ) 1/2   +dF ) 2 ))  (7)
 
 dC= 1/(4π 2   *L* (½π( L*C ) 1/2   +dF ) 2 )− C   (8)
 
     Eq. (8), gives the required dC (delta C) as a function of L, C, and dF. 
     Since dF is set to a constant, and L is constant for this discussion, dC is a function of C (frequency). 
     Notice that (4) provides F as a function of C, and dF as an independent function of dC. Since dC is now programmable (variable) by means of the tuning voltage gain stage, C determines the frequency, F, and dC determines the tuning voltage gain, dF, in an independent manner. 
     F is determined by the programmable capacitor switches which, with enough bits, gives fine digital control of center frequency. dF is determined by the programmable varactor switches, which gives continuous analog control of the frequency, ie, tuning gain. 
       FIG. 4  shows center frequency selection vs. 128 addresses (7 bits). It also shows cmin and cmax of the varactor using 15 MHz/V programmed gain for each address. The cmin/cmax ratio, when programmed, must be low enough to obtain the tuning voltage gain at the high end of the band, and high enough to obtain the desired tuning voltage gain at the low end of the band. 
     The Gm stage is consists of N selectable complementary cross coupled CMOS inverters used to offset resistive losses found in the LC tank, and is depicted in  FIG. 7 . The inverters, if operating at the right power (gm setting in this case), deliver a sinusoidal waveform. The number of individual Gm cells is determined by the total bandwidth and tank loss of the VCO. 
     When any cell is selected, the cross coupled NMOS devices are pulled to ground by the select NMOS, and the cross coupled PMOS devices are pulled to a intermediate voltage created by the active filter R 1 , C 1 , M 9 . This filter isolates supply noise from the VCO. When deselected, the cross coupled transistors are left connected to the tank, but with no path to either rail, which consequently has little affect on the selected frequency, since the device parasitics are not removed from the tank. The number of stages selected to be active is determined by performance criteria, such as noise or power, and start-up requirements, due to the load. In this case, there is no binary weighting. 
     More particularly, the inverter stages INV 1 , INV 2 , etc. are connected in common with CZ as the input signal and C as the output signal. Separate enable signals ENABLE_ 1 , ENABLE_ 2 , etc. are provided for each inverter and determine whether a particular inverter will be connected or will be disconnected (floating). Taking as an example the first inverter INV 1 , the inverter itself is formed by PMOS transistors M 11 , M 12  and NMOS transistors M 13 , M 14  connected in the same manner as previously described in relation to the oscillator  100  of  FIG. 1 . The voltage VDD is applied through a PMOS transistor M 9  to the sources of the PMOS transistors M 11  and M 12 . A voltage VSS (ground) is applied through an NMOS transistor M 15  to the sources of the NMOS transistors M 13 , M 14 . An enable signal ENABLE_ 1  is applied to a PMOS transistor M 7  and an NMOS transistor M 8  connected to form an inverter, an output signal of which is connected to the gate of the transistor M 9 . The enable signal is also inverted by an inverter IN and is applied in its inverted form to the NMOS transistor M 15 . When the enable signal is asserted, the NMOS transistor M 9  and the NMOS transistor M 15  are both caused to conduct, thereby connecting the inverter INV 1  to its supply voltages. When the enable signal is deasserted, both transistors are rendered non-conducting, removing the supply voltages from the inverter INV 1 . 
     As described in the foregoing description, a VCO, preferably wideband VCO, is provided that achieves independent control of critical VCO parameters including center frequency, tuning voltage gain and output drive strength (Gm). Incorporation of the VCO into PLLs or other systems is simplified in that operation of the VCO is readily optimized to achieve system design goals. 
     It will be appreciated by those of ordinary skill in the art that the invention can be embodied in other specific forms without departing from the spirit or essential character thereof. The foregoing description is therefore intended in all respects to be illustrative and not restrictive. The scope of the invention is indicated by the appended claims rather than the foregoing description, and all changes that come within the meaning and range of equivalents thereof are intended to be embraced therein.