Abstract:
An improved circuit and device uses a dual primary winding transformer, where the primary windings are connected in series, and where each winding has a center tapped ground. A series of four switches are attached to the ends of the two primary windings to sequentially feed the primary windings. The switches are controlled by a pair of oscillators, the first two switches on one primary winding controlled by a first oscillator, and the second two switches of the other primary winding controlled by a second oscillator. Both the first and second oscillator are controlled by a phase displacement controller which can vary the relative timing of the two oscillators to control the transformer output. A zero phase angle delivers maximum power, while a 180° phase displacement delivers zero power, enabling precise control across the zero to full power spectrum. A series of feedback controls measure the current and voltage of the output of a rectifier bridge to even more closely control the power output. A dual inverter utilizes a single transformer for light weight, efficient operation.

Description:
This is a continuation-in-part of U.S. patent application No. 08/991,981 filed Dec. 17, 1997. 
    
    
     FIELD OF THE INVENTION 
     This invention relates to a circuit and device for creating a low voltage and high current electrical power source which may be used for fusing and/or welding two or more metallic surfaces which come into contact against each other. The invention particularly relates to a circuit which eliminates low efficiency associated with the presence of both switching and saturation losses, and allows for a smaller, lighter transformer advantageous in the automated welding field. 
     BACKGROUND OF THE INVENTION 
     In a conventional &#34;H&#34; configuration alternating current power source, a direct current source is applied to a pair of parallel legs, each of which has a pair of series connected switches. A first inductor of a transformer connects the two parallel legs at a point between each pair of series connected switches. The second inductor, the transformer secondary, supplies output power based upon the opening and closing of the switches in each of the series legs. As one switch in one of the series combination upstream of the first inductor feed is closed, the other switch in the series combination is opened, while in the other series combination, the switch upstream of the first inductor connection is open, and the switch downstream switch in the series combination is closed. Once the current is allowed to flow in this configuration, the states of the switches are changed as quickly as possible to cause current to flow. 
     The problems with this design includes the switching loss, combined with the saturation loss and the current density inefficiency. In order for the switches to operate properly, all of the switches as a practical matter must be open for a short moment before any two can be closed. This is due to (1) making sure that both switches in a parallel leg are not closed at the same time which would cause a short, and (2) to make sure that the silicon controlled rectifier (SCR) switches, if used, have an opportunity to reset. The second, more important source of inefficiency is saturation loss. The losses are proportional to the product of the saturation voltage and the saturation current. The use of a single primary winding limits the output of the transformer and requires control to be limited to the temporal domain. 
     The control is based upon duty cycle, or the percentage of on time to the total time. Control of the transformer output is based upon a further limitation of the on time, from the upper limit which was already limited to give the switches a chance to clear, etc. Good lower limit control cannot be achieved efficiently since the time for current to begin to flow is not instantaneous. Thus where the pulse is controlled to occupy ten percent of the total allowable time, a ten percent power output will not occur since a significant portion of the pulse time will be spent simply enabling the pulse to rise to an acceptable point. As such, precise control cannot be had based upon percentage of on time, especially at lower percentages of the duty cycle. As the frequency of operation increases, these types of losses become even more severe. 
     These inefficiencies become even more unwieldy when used in a device for high frequency welding where a significant magnitude output current is generated. Welding and fusing operations require a relatively low voltage and hi current electrical power source for the purpose of fusing or welding two or more metallic surfaces which may be forced against each other under pressure while electric current is passed through the contact junction. The electrical current applied creates heat, which changes the molecular structure of the metallic surfaces in contact with each other and forms a metallic bond with sufficient structural holding strength at the point of contact. Precise control of heat volume and heat time at predetermined speed is required to achieve precise fusing of metals with equal and repetitive weld strength. 
     In the field of robotic welding, the current source needs to be as close as possible to the welding area for greatest efficiency and to reduce additional heat load from current flowing through cables, especially since the current is so high. The output transformer may be mounted on the robotic arm. To achieve sufficient current with the limitations outlined above, the transformer would have to be unduly large. Carrying a large, heavy transformer or current generator on the robotic arm which moves around the work piece consumes even more energy and requires a robotic arm of increased strength and which is slower, thus causing lower productivity on the line. 
     Good power transfer should be enabled in all three of high frequency, intermediate frequency and low frequency ranges. Transformers which have the size for significant power output cannot operate at high frequencies with any utilizable efficiency. The natural impedance of a large transformer would not enable it to complete its cycle at high frequency. Most high frequency welding is currently done at a relatively low frequency to enable delivery of sufficient power. It would be helpful to raise the potential operating frequencies to as high as possible to enable welding with greater control of the hot spots and more efficient application of energy to the weld point. This should be possible with an output of from about 1000 amps to about 100,000 amps under ideal conditions. 
     What is therefore needed is a circuit and device which enables high current at high frequency to be generated. The needed method should include conversion of the standard AC power source available in most industrial facilities, to a high-speed controllable voltage and current electrical power source. The power source may be programmed to produce a modulated heat by means of modulating the electrical current volume. Even more importantly, the needed circuit and device should provide itself with the capability for monitoring, and feedback to provide precise voltage, or current, or power to the fusing or welding electrodes. 
     SUMMARY OF THE INVENTION 
     An improved circuit and device uses a dual primary winding transformer having a pair of primary windings, and where each winding has a center tapped ground. A series of four switches are attached to the ends of the two primary windings to sequentially feed the primary windings. The switches are controlled by a pair of oscillators, the first two switches connected to the ends of a first primary winding controlled by a first oscillator, and the second two switches connected to the ends of a second primary winding controlled by a second oscillator. The secondary winding is divided into a pair of series connected windings with the first winding associated with the first primary and having a first core, and the second of the secondary windings associated with the second primary winding and having a second core; in essence using two separate transformers whose secondaries are series linked. Both the first and second oscillator are controlled by a phase displacement controller which can vary the relative timing of the two oscillators to control the transformer output. A zero relative displacement phase angle delivers maximum power, while a 180° relative displacement phase angle delivers zero power, given the orientation of the circuitry shown. The use of phase control enables precise control across the zero to full power spectrum. A series of feedback controls measure the current and voltage of the output of a rectifier bridge to even more closely control the power output. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The invention, its configuration, construction, and operation will be best further described in the following detailed description, taken in conjunction with the accompanying drawings in which: 
     FIG. 1 is an overall schematic of the voltage and current bearing components of the circuit of the invention; 
     FIG. 2 illustrates the control layout of the circuit and which is keyed to FIG. 1; 
     FIG. 3 illustrates graphically the states of the switches and outputs of the circuit of FIG. 1 at a phase displacement of 180°, 0° &amp; 90°; 
     FIG. 4 illustrates graphically the states of the switches and outputs of the circuit of FIG. 1 at a phase difference of 0°; 
     FIG. 5 illustrates graphically the states of the switches and outputs of the circuit of FIG. 1 at a phase difference of 90°; and 
     FIG. 6 is an overall schematic illustrating a variation of the voltage and current components of the circuit of FIG. 1 where a single transformer is utilized; 
     FIG. 7 is an overall schematic illustrating a further variation of the voltage and current components of the circuit of FIG. 1 where a single transformer is utilized, and where the primary windings are not center tapped, and where exact switching control is obtained; 
     FIG. 8 is a timing diagram where the primary windings of the transformer of FIG. 7 are zero degrees out of phase to produce full output; 
     FIG. 9 is a timing diagram where the primary windings of the transformer of FIG. 7 are ninety degrees out of phase to produce half output; 
     FIG. 10 is a timing diagram where the primary windings of the transformer of FIG. 7 are one hundred eighty degrees out of phase to produce a zero output; 
     FIG. 11 is a schematic view of the winding placement of the transformer used in the circuit of FIG. 7; 
     FIG. 12 is a diagram illustrating the orientation of the four primary windings for a high voltage input; the four primary windings for a low voltage input, both with respect to the orientation of the secondary windings. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     The description and operation of the invention will be best initiated with reference to FIG. 1. At the upper left side of FIG. 1, a three phase source 11 supplies electricity through three lines 13, 15, &amp; 17 to a three phase rectifier bridge 21. The three phase rectifier bridge 21 has a pair of direct current outputs, including a first line 23 and a second line 25. A filtering capacitor C1 is connected between first direct current line 23 and second direct current line 25 to short any alternating current components still present, and provide a low impedance to an instantaneous current demand. 
     A pair of transformers are shown and labeled T1 and T2. The T1 transformer has a center tapped primary coil 31 and a secondary coil 33. The center tap of the primary coil 31 is connected to the first direct current line 23. The ends of the primary coil 31 are connected to a pair of switches, one end of coil 31 is connected to one terminal of a first switch SSSW1, the other end of switch SSSW1 connected to the second direct current line 25. The other end of coil 31 is connected to one terminal of a first switch SSSW2, the other end of switch SSSW2 is connected to the second direct current line 25. 
     Transformer T2 transformer has a center tapped primary coil 35 and a secondary coil 37. The center tap of the primary coil 35 is connected to the first direct current line 23. The ends of the primary coil 35 are connected to a pair of switches, one end of coil 35 is connected to one terminal of a first switch SSSW3, the other end of switch SSSW3 connected to the second direct current line 25. The other end of coil 35 is connected to one terminal of a first switch SSSW4, the other end of switch SSSW4 is connected to the second direct current line 25. 
     The transformer T1 secondary coil 33 has a first end connected into a rectifier bridge 41 and a second end. The transformer T2 secondary coil 35 has a first end connected into the second end of secondary coil 33 of transformer T1, and a second end connected into the rectifier bridge 41. 
     The output of the rectifier bridge 41 includes a first low voltage high current line 45 and a second low voltage high current line 47. Lines 45 and 47 have a filter in the form of a series connected inductor L1 in the line 45 and a capacitor C2 connected between the lines 45 and 47. Downstream of the filter capacitor C2 is a current loop 49 surrounding the line 47 for measuring the current in line 47. Current loop 49 has a pair of leads 50 extending away from the current loop 49. A pair of voltage test leads 51 and 53 are seen, with test lead 51 connected to line 45 and test lead 53 connected to line 47. The ends of lines 45 and 47 are triangularly shaped to symbolize their application to an article to be welded or fused, and are seen as welding or fusing electrodes 55 and 57. 
     Each of the switches SSSW1, SSSW2, SSSW3, &amp; SSSW4 have an actuation line, namely a line 61, 63, 65, and 67 respectively. The actuation lines 61, 63, 65, and 67 control the opening and closure of the switches SSSW1, SSSW2, SSSW3, &amp; SSSW4. 
     Referring to FIG. 2, it can be seen that the lines 61 &amp; 63 are connected to an oscillator 71, while lines 65 &amp; 67 are connected to an oscillator 73. Both of the oscillators 71 and 73 are connected to and independently controlled by a phase controller 75. Into the phase controller is the beginning of a control system which controls the operation of the oscillators 71 and 73, and their relative timing of operation, based upon the current and voltage levels requested, and by adjusting for current and voltage feedbacks from the current loop 49 and the pair of voltage test leads 51 and 53. It is understood that many other methods and configuration of control are possible, and that this is just one configuration. 
     In the control configuration shown, the phase controller 75 is directly connected to a programmed comparator 81. Programmed comparator 81 has a first input from a programmable controller 83 and a second input from a voltage current power feedback block 85. The programmable controller 83 is also connected back into the voltage current power feedback block 85. 
     The voltage current power feedback block 85 is connected into a voltage and current multiplier block 87, a current feedback signal conditioner 89, and a voltage feedback signal conditioner 91. The voltage and current multiplier block 87 receives inputs from both the current feedback signal conditioner 89, and the voltage feedback signal conditioner 91. The current feedback signal conditioner 89 is connected to the leads 50 of the current loop 49. The voltage feedback signal conditioner 91 is connected to the voltage test leads 51 &amp; 53. 
     Referring to FIGS. 3, 4, &amp; 5, a series of three diagrams illustrating a 180°, 0° and 90° phase displacement relationship are shown together in order to utilize a common row label, which is shown to the right of FIG. 5. 
     Referring to FIG. 3, a diagram illustrating a 180° phase displacement relationship for the oscillators 71 and 73 which corresponds to voltages across the switches SSSW1, SSSW2, SSSW3, and SSSW4, the secondary windings 37 and 33, and the voltage across the output of the rectifier bridge 41 as seen across capacitor C2, is shown. 
     The OSCILLATOR 71 (labeled #1) and OSCILLATOR 73 (labeled #2) are two symmetrical square wave generators operating at the same frequency. Their purpose is to turn `on` and `off` the 4 solid state switches SSSW1, SSSW2, SSSW3, and SSSW4, using the lines 61, 63, 65, &amp; 67. The solid state switches SSSW1, SSSW2, SSSW3, and SSSW4 are turned on and off alternately at 50% duty cycle. When the odd number switches are on, the even number switches are off and vice versa. It should be noted on FIG. 1 that the winding polarity of transformer T1 is such that transformer magnetic flux flows in one direction when SSSW1 is on and SSSW2 is off, and in the other direction when SSSW2 is on and SSSW1 is off. The same is true for the windings of transformer T2, and its associated switches SSSW3 and SSSW4. The illustration of the voltage across the solid state switches can be seen in FIG. 3. 
     The vertical scale of FIG. 3 is a relative phase scale and is marked at 90° intervals beginning at 0°, with the 360° mark labeled as 0° again. As is seen, the vertical section time period is useful for illustrating time events, as well as relative angular phase displacement. The changes taking place across the phase progression occur at 180° time intervals, for example. During the first 180° period, the voltage across SSSW1 is high (open condition), SSSW2 is low (closed condition), SSSW3 is low (closed condition), SSSW4 is high (open condition). This urges the currents in the series connected secondary coils 33 and 37 in opposite directions. Here, current in the coil 31 is taken as producing a positive voltage (seen as the waveform labeled VOLTAGE ACROSS T1 SEC.), while current in the coil 35 is taken as producing a negative voltage (seen as the waveform labeled VOLTAGE ACROSS T1 SEC.). This results in a net zero flow of power both into and out of the bridge rectifier 41. This is also seen by visually comparing the current flow tendency across the secondary windings 37 and 33, and noting that they are opposite, and algebraically add to zero. The algebraic sum is seen in the waveform labeled VOLTAGE ACROSS RECTIFIED OUTPUT. The solid state switches transition spikes shown are filtered out if need be, and of no operational consequence. The bottom-most waveform is the AVERAGE FILTERED VOLTAGE ACROSS POWER SUPPLY OUTPUT and is shown to have a resultant voltage of 0 volts, and which appears at the capacitor C2. 
     Note that the voltage scale used for the voltages across the switches SSSW1, SSSW2, SSSW3, and SSSW4 is shown to be 700 volts, and that this magnitude can vary from case to case as the design dictates, 700 volts chosen for convenience of understanding. This results in a voltage output at C2 which is zero, due to the algebraic addition of the voltage seen across the T2 secondary at -5 volts, to the voltage seen across the T1 secondary at +5 volts, and is also shown by the 0 V line at the bottom diagrams. The same result occurs during the time which elapses between the 180° and 0° time periods. 
     Referring to FIG. 4, the second Figure in the series of three, the relationship of 0° phase difference is seen, and which will produce maximum power output. Referring to the first 180° of time, during the first set of states, the voltage across SSSW1 is high (open condition), SSSW2 is low (closed condition), SSSW3 is high (open condition), SSSW4 is low (closed condition). This urges the currents in the series connected secondary coils 33 and 37 in the same direction resulting in a double net flow of power both into the bridge rectifier 41. This is also seen by comparing the waveforms VOLTAGE ACROSS T2 SEC. with the VOLTAGE ACROSS T1 SEC and remembering that they algebraically add. Visually, it can be seen that the current produced across the secondary windings 37 and 33 is of the same polarity and double the voltage going into the rectifier bridge 41. With the algebraic sum of the two secondary windings T1 and T2 being added, the thus rectified and filtered output is at maximum attainable voltage. This results in a voltage output at C2 which is 10 volts, and is also shown by the 10 V line at the bottom diagrams. 
     The waveforms shown in FIG. 5 represents the 90° phase relationship, for half the maximum power output. Note that the first two wave forms representing SSSW1 and SSSW2 have a fixed relationship, as must be the case to avoid a short. The same is true for the states of SSSW3 and SSSW4. However note that the waveforms for SSSW3 and SSSW4 are shifted half way to the one side with respect to SSSW1 and SSSW2. This causes them to be in phase for half of the time that they assume a fixed state and out of phase for the other half of the time in which they are in their fixed state. This produces an instantaneous 10 volts output for one quarter of the time, an instantaneous 0 volt output for one quarter of the time, and an instantaneous 5 volt output half of the time. The average for these voltage magnitudes over time is 5 volts. The averaging displayed in FIG. 3 is the filtered output with a high resistive load. The used output voltage is the equivalent RMS average as it appears across the electrode shorting the metals to be welded or fused. This averages to 5 volts, and indeed the averaging may physically occur through C2 if it is large enough, or if the time periods are short enough. It is understood that FIGS. 3, 4, &amp; 5 show the extreme relationships, and that phase relationships producing an amount of voltage from 0% to 100% is contemplated, based upon the circuit shown. 
     Referring to FIG. 6, a variation on some of the circuitry seen in FIG. 1 is shown, and where the circuit components are the same and perform the same function as the circuitry of FIG. 1, they are labeled the same. The FIG. 6 circuit combines two inverters into a single transformer. The circuit begins downstream of the three phase rectifier bridge 21 which is not shown, where the lines 23 and 25 receive a high voltage direct current source, such as about 340 volts D.C. The switches SSSW1, SSSW2, SSSW3, &amp; SSSW4 are not shown as connected to their controlling oscillators 71 &amp; 75, only for simplicity, and the operation of the circuit will be the same as for FIG. 1 and will require such connection. 
     Here, a single transformer T3 is provided which contains the center tapped primary windings 31 and 35, as were shown in FIG. 1, but magnetically linked through a single core. The phase of the two inverters in the primary coils 31 and 35 are controlled as was described for FIGS. 3-5, with the two primary coils 31 and 35 of a pair of input inverters having the ability to phase sum in order to achieve control. The windings 31 and 35, however, in a single transformer T3 will be wound on opposite sides of the same transformer core. Transformer T3 has a single center tapped secondary winding 95. One end of winding 95 is connected to the current input end of a diode D1, while the other end of winding 95 is connected to the current input end of a diode D2. The current output ends of the diodes D1 and D2 are connected together and become the first low voltage high current line 45 seen in FIG. 1. The single center tapped secondary winding 95 becomes the second low voltage high current line 47 seen in FIG. 1. Lines 45 and 47 continue exactly as seen in FIG. 1 with the filter in the form of a series connected inductor L1 in the line 45 and a capacitor C2 connected between the lines 45 and 47. The circuit of FIG. 4 provides more efficient rectification due to the reduction of current and resistance (IR) losses and transformer core losses. 
     An alternative embodiment using precise timing is seen in FIG. 7 as an overall schematic device 101. The details of the three phase rectifier bridge 21 seen in FIG. 1, are also schematically illustrated also. A three phase input power source is represented by lines L1, L2, and L3. Each is connected between a current input of associated one of diodes D11, D13, and D 15, and a current output of associated ones of diodes D17, D19 &amp; D21. The current output sides of diodes D11, D13, and D 15 are connected to one end of a capacitor C11, while the current input sides of diodes D17, D19, and D 21 are connected to the other end of capacitor C11. 
     One side of Capacitor C11 is connected to a series of switches, all having a (+) designation, in order to simplify later discussion, and including SSSW1(+), SSSW2(+), SSSW3(+), &amp; SSSW4(+). Note that the (+) designation corresponds with current flow into the switches taking on a positive orientation. The other side of Capacitor C11 is connected to a series of switches, all having a (-) designation, in order to simplify later discussion, and including SSSW1(-), SSSW2(-), SSSW3(-), &amp; SSSW4(-). Note that the (-) designation corresponds with current flow our of the switches, back toward diodes D17, D19, &amp; D21, also taking on a positive orientation. Because this convention is used, the voltage drop over each associated pair of switches will be the same and will be easier to track on a timing diagram. 
     A winding A is connected in series with a winding B. The A and B series windings can be connected in parallel, as will be shown for another configuration. In FIG. 7, switches SSSW1(+) and SSSW1(-) are connected to the end of the A and B series at the A winding. Switches SSSW2(+) and SSSW2(-) are connected to the other end of the A and B series at the B winding. Similarly, switches SSSW3(+) and SSSW3(-) are connected to the end of the C and D series at the C winding. Switches SSSW4(+) and SSSW4(-) are connected to the other end of the C and D series at the D winding. Windings A, B, C, &amp; D form the primary to a transformer T11. The secondary for the transformer T11 includes a series combination of windings E and F having a center tap. 
     Winding E has its end not connected to winding F or the center tap, connected through a diode CR1 and then to a positive end of a welding electrode 103. The center tap is connected directly to a negative end of a welding electrode 105. Winding F has its end not connected to winding E or the center tap, connected through a diode CR2 and then to a positive end of welding electrode 103. In either of the configurations of device 101 to be shown, the electrical potential developed between electrodes 103 and 105 is preferably maximum at about ten volts direct current, and rated at about twenty thousand amps at shorted output. 
     The switches SSSW1(+), SSSW2(+), SSSW3(+), &amp; SSSW4(+), SSSW1(-), SSSW2(-), SSSW3(-), &amp; SSSW4(-) are connected to oscillators 71 and 73, which are the same oscillators of the control circuitry shown in FIG. 2. The switches SSSW1(+), SSSW2(+), SSSW3(+), &amp; SSSW4(+), SSSW1(-), SSSW2(-), SSSW3(-), &amp; SSSW4(-) are connected in pairs to operate in simultaneous combination to force current through the A and B coils in one direction and then in the opposite direction, and to force current through the C and D coils in one direction and then in an opposite direction. The switch pairs SSSW1(+) and SSSW2(-) are connected to actuation line 61, switch pairs SSSW1(-) and SSSW2(+) are connected to actuation line 63, switch pairs SSSW3(+), and SSSW4(-) are connected to actuation line 65, and switch pairs SSSW3(-), and SSSW4(+) are connected to actuation line 67. The oscillators 71 and 73 in this case are expected to provide exact timing control to insure that the 0.002 millisecond time spacing between switch actuations can be adequately controlled. Again, controller 75 controls the phase difference between the operation of the first two switch pairs SSSW1(+)/ SSSW2(-) &amp; SSSW1(-)/ SSSW2(+), with respect to the timing of the operation of the second two switch pairs SSSW3(+)/SSSW4(-) &amp; SSSW3(-)/SSSW4(+). 
     The operation of the circuit of FIG. 7 is best shown with respect to the timing diagram of FIG. 8. The entire two periods shown is preferably about 0.1 microseconds, equal to a full period for each switch of about 0.05 microseconds. Two full three hundred sixty degree periods are shown so that the phase difference between the periods can be shown in a shifted relationship and be more readily understood. 
     For the full cycle of 0.05 microseconds the switch activation time is 0.046 microseconds, for a 92% duty cycle. The time period between change of state of the switches is 0.002 microseconds. For each full period, this occurs twice and represents an 8% off duty time. The voltage of the switches between on and off states is expected to be in transition, although the transition is desired to take place as rapidly as possible. The phase setting of FIG. 8 is expected to produce maximum output. 
     Referring to FIG. 9, all of the time spacing of the first two switch pairs SSSW1(+)/ SSSW2(-) &amp; SSSW1(-)/ SSSW2(+) is the same, as are the time spacing of the second two switch pairs SSSW3(+)/SSSW4(-) &amp; SSSW3(-)/SSSW4(+), but the relative spacing between the two is displaced by ninety degrees. The phase setting of FIG. 9 is expected to produce half the maximum output. 
     Referring to FIG. 10, the relative spacing between the two is displaced by one hundred and eighty degrees. The phase setting of FIG. 10 is expected to produce a zero output. 
     Referring to FIG. 11, a schematic of the transformer T11 includes a first core 111 and a second core 113. The coils are arranged along a common inside core, which includes about half of the elongate length of the cores 111 and 113, in the order AEBFCEDF. Thus the E and F coils appear twice on the core and are interstitially placed between the A, B, C, and D coils. 
     Referring to FIG. 12, a configuration 131 is shown for the primary coils A, B, C, &amp; D, which is reflective of the configuration shown in FIG. 7. A configuration 133 is shown for the Secondary coils E and F. Note that E is actually two coils connected in parallel, which is why the E and F coils in FIG. 11 appear in two different locations. The configuration 131 when used in conjunction with the secondary configuration 133 is the step down configuration to be used for a 480 volt AC primary three phase input at the lines L1, L2, and L3 of FIG. 7. 
     Also seen is a configuration 135 which can be used in conjunction with secondary configuration 133 in which the transformer T11 of FIG. 11 and its physical layout is still possible. The difference is that configuration 135 includes coils A and B which are connected in parallel, as are coils C and D. In this configuration, switch pair SSSW1(+)/ SSSW2(-) will be connected to one end of the parallel combination of Coils A &amp; B, while switch pair SSSW1(-)/ SSSW2(+) will be connected to the other end of the parallel combination of Coils A &amp; B. Likewise, switch pair SSSW3(+)/SSSW4(-) will be connected to one end of the parallel combination of Coils C &amp; D, while switch pair SSSW3(-)/SSSW4(+) will be connected to the other end of the parallel combination of Coils C &amp; D. 
     While the present invention has been described in terms of a circuit to be used in welding and the like, one skilled in the art will realize that the structure and techniques of the present invention can be applied to many similar devices. The present invention may be applied in any situation where a phase control or power averaging is to be used electrically to achieve desired electrical or electronic output. 
     Although the invention has been derived with reference to particular illustrative embodiments thereof, many changes and modifications of the invention may become apparent to those skilled in the art without departing from the spirit and scope of the invention. Therefore, included within the patent warranted hereon are all such changes and modifications as may reasonably and properly be included within the scope of this contribution to the art.