Abstract:
A double balun dipole antenna element includes a dielectric substrate having a first surface and an opposing second surface, a pair of coplanar Marchand baluns positioned in a mutually antiphase configuration on the first and second surfaces, and at least one feed line connected to the pair of Marchand baluns. A doubly polarized antenna element includes a pair of orthogonally interleaved double balun dipole antenna elements, which can be further configured into an array of such antenna elements.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This Application is a Non-Prov of Prov (35 USC 119(e)) application 60/972,422 filed on Sep. 14, 2007. 
    
    
     TECHNICAL FIELD 
     The present invention is directed to an antenna element for an ultra wideband array antenna. More particularly, the invention is directed to a double Marchand balun dipole antenna element for an antenna array. 
     BACKGROUND OF THE INVENTION 
     There has been increasing interest in coincident phase center elements for electronically steered, polarization diverse, ultra wideband array antennas in recent years. This interest has arisen from the difficulty of maintaining the axial ratio of circularly polarized beams when scanning off-axis. When the constituent linear polarizations used to form circular polarization have adjacent phase centers, an angle dependant path length difference is introduced upon scanning. With increasing bandwidth, compensating for this path difference becomes more difficult. The motivation for developing ultra wideband coincident phase center antennas is to eliminate the scan dependant path length difference associated with adjacent phase center antennas. 
     Navy ships require electronically steerable antenna arrays capable of transmitting and receiving signals with polarization diversity, including circularly polarized waves, over large instantaneous bandwidth for satellite communications, electronic warfare, and other applications. Antenna element designs include those for operation using different polarizations, including linear (vertical or horizontal) and circular. These antenna elements are typically assembled into arrays for generating or receiving a collimated, directed RF beams. 
     To obtain polarization diversity, an antenna needs to radiate two orthogonal polarizations independently. This can be done with a pair of orthogonally positioned, linearly polarized elements. If electronic steering of circularly polarized beams is desired, then the linearly polarized elements must also have coincident phase centers to avoid degradation of circularity as the beam is scanned. The polarization purity degrades further in the case of wide instantaneous bandwidth signals. 
     Ultra wideband antenna arrays frequently employ flared notch radiators. This is because flared notch radiators usually do not have a strong resonance, but rather may be viewed as smooth tapers from a confined transmission line mode to a radiating free space mode. The difficulty with flared notches is that the individual radiators require conductive contact between adjacent elements to operate correctly at low frequencies within their design bandwidths 
     Employing elements which do not require conductive contract between adjacent elements may free the designer from the difficulties of maintaining electrical contact between adjacent elements, but may introduce problem of obtaining large bandwidths from elements not known for wide bandwidth. 
     With regard to bandwidth enhancement, there are two frequency regimes to consider: the high frequency regime where in a half wavelength is less than the array cell size and the low frequency regime where in a half wavelength greater than the array cell size. The low frequency regime is more interesting for electronic beam steering applications while the high frequency regime is more interesting for fixed scan or broadside applications. In the low frequency regime a two to one bandwidth is readily attainable while maintaining simple construction methods. A four to one bandwidth is achievable with special construction techniques. The limits of the high frequency regime are encountered when interference between direct radiation from the dipoles, and reflected radiation from the ground plane behind the dipoles begins to form a null in the radiation pattern. 
     There are some trade off between low and high frequency performance. The more the bandwidth is extended in the low frequency regime, the greater the reflections seen in the high frequency regime. When the special construction techniques are employed to extend the bandwidth to 4:1 in the low frequency regime, there is very little bandwidth left in the high frequency regime. 
     A cross sectional view of a unit cell  10  in a coincident phase center array antenna with a coincident phase center flared notch element is shown in  FIG. 1 . This antenna is difficult to build because it must be constructed to maintain electrical continuity in multiple places for two interleaved perpendicular polarizations simultaneously. The first polarization is in the plane  12  of the sectional cut, and has its feed circuitry  14  exposed. The second polarization plane  16  is perpendicular to the section plane. Within a unit cell, it is important to maintain microwave electrical continuity across the slot  18  (further described below in an E-plane tee configuration). Between unit cells it is important to maintain microwave electrical continuity at the flare tips  20 . Here microwave continuity implies control of the geometry of mating conductors so that the smooth flow of microwave fields can occur. Direct current continuity simply requires that mating conductors be in contact. The double balun dipole according this intention does not require continuity between unit cells in an array which greatly simplifies construction. 
     The next design to be considered is not a coincident phase center antenna, but it is introduced to help explain later designs.  FIG. 2  shows a microstrip fed printed dipole with an integral balun by Edward and Rees, as described in U.S. Pat. No. 4,825,220, issued Apr. 25, 1989. It has a single Marchand balun  50  feeding the two arms of the dipole. It has a bandwidth 3 or 4 times greater than a traditional split sleeve dipole and it is fabricated using printed circuit techniques. 
     Before discussing the next patents/prior art, it will be useful to address microwave transmission lines. This will help to understand the shortcomings of the patents/prior art to be discussed. The classic transmission line is the parallel wire transmission line  60  shown in  FIG. 3 . The two conductors  62  are the same size, and the currents which flow on them are mirror images of one another. The difficulty with parallel wire transmission lines is that they must be protected from the rest of the world. A conductor that is allowed too close can cause reflections or permit extra modes to propagate, and dielectrics can change the characteristic impedance or introduce losses. A slot  64  is also shown in  FIG. 3  as an example of balanced transmission line. Unless the conductors  66  and  68  on each side of the slot have the same cross-section, the transmission line mode it supports is only approximately balanced. In general slots tend to radiate, so it is desirable to keep them short with respect to a wavelength. In practice most microwave transmission lines are unbalanced lines such as those shown in  FIG. 4 . One conductor  70  is made much larger than the other conductor  72 , and is often grounded. It shields the smaller conductor from the surroundings, making the circuit much easier to work with. Stripline has two ground conductors, but they are often joined with screws or plated vias, so it is similar to coaxial line. Direct connections between balanced and unbalanced transmission lines generally result in undesirable large reflections. Usually a balun circuit is inserted at the junction to enable proper flow of signals across the boundary. 
     The groundwork has been laid to consider the Wideband Phased Array Antenna and Associated Methods by Munk, Taylor, and Durham (“Munk et al.”), as described in U.S. Pat. No. 6,512,487 and in “A Wide Band, Low Profile Array of End Loaded Dipoles with Dielectric Slab Compensation,” Ben A. Munk, 2006 Antenna Applications Symp., pp. 149-165 (“Munk”), shown in  FIG. 5 . In the patent Munk et al. claims a 15:1 bandwidth, while in the paper Munk shows plots indicating a 9:1 bandwidth. Both are remarkable. Through the use of interdigital capacitors at the ends of the dipole arms, and selective use of dielectric layers, Munk and Munk et. al. have constructed a dipole array which is intrinsically well matched over an extremely wide bandwidth. Having constructed well matched dipoles, they must feed them with a balanced transmission lines. However, as I pointed out above, balanced transmission lines are undesirable because they need to be shielded. Munk et al. does not describe the particulars of the feed network in the patent. However, another patent, U.S. Pat. No. 6,483,464, Rawnick et al., describes a feed network and associated feed line organizer, shown in  FIG. 6 , which they state is suitable for the dipole array antenna patented by Munk et al. Rawnick et al. state that the feed line organizer suppresses common mode currents. Indirectly, this may be an affirmation of the difficulty noted above with balanced transmission lines. An isolated single two conductor transmission line should support only one mode of propagation. If a second two conductor transmission line is brought into close proximity of the first, the result is a four conductor transmission line which can support unwanted modes. The self-shielding of unbalanced transmission lines reduces the coupling to nearby transmission lines. In effect, Rawnick et al. has replaced two balanced transmission lines in close proximity with four unbalanced transmission lines with the feed line organizer. A 180 degree hybrid circuit is also shown in  FIG. 6 . The balanced transmission line mode of Munk et al. and Munk&#39;s dipole still needs to be converted to an unbalanced mode. 
     Typically a 180 degree hybrid has several quarter wavelength sections. This is at some intermediate frequency, not the highest frequency. However the space available is a square one half wavelength on a side, and this is at the highest frequency. The size of the circuit can be reduced by the use of high dielectric constant materials, but only to a point. Practical circuit processing techniques limit how small features of a circuit can be made. Munk&#39;s design is very clever. However the dipole according to this invention has the advantage of providing unbalanced transmission line modes right at the terminals of the antenna. The conversion from a balanced mode to an unbalanced mode is implemented more efficiently in less space with a double Marchand balun dipole than with a balanced dipole, feed line organizer, and 180 degree hybrid. Munk notes that his design is capable of dual polarized operation, but he does not mention coincidence of phase centers. It is therefore desirable to provide a dipole antenna without these deficiencies. 
     BRIEF SUMMARY OF THE INVENTION 
     According to the invention, a double balun dipole antenna element includes a dielectric substrate having a first surface and an opposing second surface, a pair of coplanar Marchand baluns positioned in a mutually antiphase configuration on the first and second surfaces, and at least one feed line connected to the pair of Marchand baluns. In the microstrip embodiment, the dipole is positioned on the surface opposite the feed line. In the stripline embodiment, the feed line is positioned as is described below. A doubly polarized antenna element includes a pair of orthogonally interleaved double balun dipole antenna elements, which can be further configured into an array of such antenna elements. 
     With this invention, dipoles are employed as radiating elements. A technique using two Marchand baluns, one for each arm of the dipole, is introduced for enhancing the bandwidth of dipole radiators. Herein the Marchand baluns server two purposes: (1) converting the balanced field mode of the dipole to the unbalanced mode of the transmission line, and (2) matching the capacitive loading of adjacent dipoles. 
     The advantages of the double Marchand balun dipole of the invention vary depending on the particular embodiment. However, an advantage common to all embodiments is that it operates over a considerably wider bandwidth than most dipole antennas. 
     The invention, unlike a dual-polarized microstrip notch antenna, does not require electrical continuity between contiguous elements, greatly simplifying and reducing the costs of its construction. 
     The insight gained from designing ultra wideband coincident phase center elements can be used to redesign more narrow band elements to have coincident phase centers. Of course the unique characteristics of different elements must be accounted for with any new design. This method is applied to dipoles with this invention. At the same time, some emphasis will be placed on keeping the resulting design simple. 
     Two double Marchand balun dipoles can be arranged so that they are mutually perpendicular to each other and yet share a common physical center. In this configuration they can be used for coincident phase center applications. Coincident phase center applications are likely to be associated with electronic beam steering. Electronic beam steering implies element spacing of approximately a half wavelength or less—the low frequency regime. The bandwidth of the double Marchand can be expanded in the low frequency regime by making the arms of the dipoles longer, the limit being half the cell size of the array. A two to one bandwidth is easily attainable. A four to one bandwidth is attainable by using capacitors between dipoles as Munk did. In this embodiment, the advantage of the double Marchand balun dipole is that its ports are unbalanced transmission lines. 
     The double Marchand balun dipole can be used in wideband single polarized applications also. The bandwidth can be increased in the low frequency regime for single polarization applications the same way it is for coincident phase center applications. I do not believe ordinary split sleeve balun dipoles are susceptible to the same type of bandwidth enhancement. I have observed mutual coupling induced bandwidth enhancement with the Edward&#39;s dipole, but I believe the double Marchand balun dipole is susceptible to more enhancement. 
     The double Marchand balun can be used as an array element in the high frequency regime also. In this frequency range it is more likely to be useful for fixed scan or broadside applications. 
     The double Marchand balun dipole has usefulness in single or isolated antenna applications. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a coincident phase center array antenna with a coincident phase center flared notch element; 
         FIG. 2  is a microstrip fed printed dipole with an integral balun; 
         FIG. 3  illustrates parallel wire and slot balanced transmission lines; 
         FIG. 4  illustrates coax, stripline, and microstrip unbalanced transmission lines; 
         FIG. 5  illustrates array antenna configurations; 
         FIG. 6  illustrates a feed network and associated feed line organizer; 
         FIGS. 7  A-B respectively illustrate a regular and an E-plane transmission line tee; 
         FIG. 8  is a schematic illustration of a double Marchand balun feed according to the invention; 
         FIG. 9  is a schematic illustration of the balun and slotline portions of  FIG. 8 ; 
         FIG. 10  is alternative representation of the double balun dipole of  FIG. 8 ; 
         FIG. 11  is a schematic illustration of double Marchand balun dipole radiators positioned end to end or in linear arrays as integrally constructed on circuit boards according to the invention; 
         FIG. 12  is an equivalent circuit representation of the configuration shown in  FIG. 9 ; 
         FIG. 13  is exploded view of a stripline dipole comparable to the microstrip dipole shown in  FIGS. 11 and 8  according to the invention; 
         FIG. 14  is the calculated response for the dipole of  FIG. 13 ; 
         FIGS. 15A-B  are double Marchand balun dipoles at the end or edge of an array according to the invention; 
         FIG. 16  is the calculated response for the element without conducting end tabs and the calculated response for the element with conducting end tabs according to the invention; 
         FIGS. 17A-B  are schematic illustrations of an assembly of two perpendicular stripline dipoles according to the invention; 
         FIGS. 18A-B  are microstrip double balun dipoles in a coincident phase center configuration according to the invention; 
         FIG. 19  shows construction details of a double Marchand balun dipole in an array environment according to the invention; 
         FIG. 20  illustrates the use of end capacitors in an array according to the invention; and 
         FIG. 21  shows simulation results of the array of  FIG. 20 . 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Before going into detail on operation and construction of the double Marchand balun dipole of the invention, it may be helpful to elaborate on the difference between a regular tee and E-plane tee. The term “E-plane tee” is borrowed from waveguide usage. Both type of transmission line tees are shown in  FIGS. 7A-B . The tee in  FIG. 7A  is a regular tee. The electric field has the same polarity at both outputs. The tee in  FIG. 7B  is an E-plane tee. The electric field at the outputs has opposite polarity. Both tees can be implemented with balanced transmission line topology. The E-plane tee cannot be implemented with unbalanced transmission line topology. This is because the roles the two conductors play, as “shield” and “center conductor” would have to be reversed at the two output arms. 
     Referring back to  FIG. 4 , a double balun dipole can be implemented with either microstrip or stripline circuit methods. A double balun dipole  80  implemented with microstrip techniques is shown in  FIG. 8  (in this drawing, the top, dielectric substrate is shown in phantom so that both the top and bottom sides may be shown in the one figure). It consists on the bottom side of a conducting dipole  82  and on the opposing top side a conducting feed circuit  84  printed on a dielectric substrate  86 . The feed circuit consists of a tee  88 , feed lines  90  and  92 , and open circuited stubs  94  and  96 . Nominally the feed lines  90  and  92  have the same electrical length although they may follow different contours. The same applies to the open circuited stubs  94  and  96 . The dipole  82  is bifurcated at its midpoint by slotline  98  which leads to E-plane tee  100 . E-plane tee  100  feeds short circuited slotline stubs  102  and  104  (thus positioned on the bottom, conducting dipole  82  side, i.e. the side opposite the top side of stubs  94  and  96 ) in anti-phase. Nominally the short circuited stubs  102  and  104  have the same electrical length. Feed lines  90  and  92  cross short circuited stubs  102  and  104 , in opposite directions, at symmetrical distances, which are kept as short as possible, from E-plane tee  100 . Immediately upon crossing short circuit stubs  102  and  104 , feed lines  90  and  92  transition into open circuit stubs  94  and  96  which may have the same or different widths as feed lines  90  and  92 . Collectively open circuit stub  94  and short circuit stub  102  form a Marchand balun  106 , and open circuit stub  96  and short circuit stub  104  form a Marchand balun  108 . In essence, the single Marchand balun of Edward&#39;s design,  FIG. 5 , has been replaced with two counter-phased Marchand baluns joined with an E-plane tee. 
     The balun and slotline portions of  FIG. 8  are shown schematically in  FIG. 9 . In this schematic unbalanced transmission lines are shown as small round conductors adjacent to flat conductors while balanced transmission lines are shown as to two adjacent flat conductors. Unbalanced feed lines  90  and  92 , with signals which are in phase, feed Marchand baluns  106  and  108  respectively. Marchand balun  106  consists of open circuited stub  94  in series with input  90  and short circuited stub  102  in parallel with output  120 . Similarly Marchand balun  108  consists of open circuit stub  96  in series with input  92  and short circuited stub  104  in parallel with output  122 . Marchand balun  106  and  108  are oriented so as to feed balanced transmission lines  120  and  122  with signals which are oppositely phased. These signals are combined by E-plane tee  100  and passed to balanced transmission line  98 . Optimally, balanced transmission lines  120  and  122  are kept as short as possible to minimize the distance between the E-plane tee and the Marchand baluns. Note that the short balanced lines  120  and  122  were not distinguished from stubs  102  and  104  in  FIG. 8 . They were drawn with the same width as stubs  102  and  104 , and specifically identifying them might have been confusing. 
     An alternative representation of the double balun dipole, useful for further explaining its operation is shown in  FIG. 10 . The left and right halves of dipole and feed circuit are shown as reflections of one another in mirror plane  130 . The dipole  12 , slot  98 , E-plane tee  100 , feed line  90 , open circuited stub  94  and short circuited stubs  102  and  104  are the same as shown in  FIG. 8 . The feed line  132  and open circuited stub,  134 , on the opposite side are exact mirror images of the first. The direction in which feed line  132  crosses short circuited stub  104  is the same as that in which feed line  90  crosses short circuited stub  102  so the reversal in phase between the two sides of the dipole is obtained with a 180° hybrid circuit  136 . 
     Array Embodiment 
     Double Marchand balun dipole radiators may be integrally constructed on circuit boards and positioned end to end or constructed in linear arrays on circuit boards as shown in  FIG. 11 . Bandwidth enhancement is obtained by making the arms of adjacent dipoles  140  and  142  approach each other closely taking advantage of the capacitive coupling noted by Munk. As the capacitive loading increased, the reactive fields of the dipole change, and the lengths of the open and short circuited stubs in the Marchand baluns may be changed to compensate. This can be seen by redrawing the schematic in  FIG. 9  to emphasize the circuit parameters rather than the transmission line characteristics of the double Marchand balun. Referring to  FIG. 12 , open circuited stubs  94  and  96  are represented by capacitors with impedance −jZ oc  COT(β oc l oc ) and short circuited stubs  102  and  104  are represented by inductors with impedance jZ sc  TAN(β sc l sc ) where
         Z oc , is the characteristic impedance of the open circuited stubs,   β oc  is the propagation constant of the open circuited stubs,   l oc  is the length of the open circuited stubs,   Z sc  is the characteristic impedance of the short circuited stubs,   β sc  is the propagation constant of the short circuited stubs, and   l sc  is the length of the short circuited stubs.       

     The end loading is represented by impedances Z end  which is the capacitive end loading already mentioned transformed by the length of the monopole  150 . The schematic in  FIG. 12  is not complete enough to design a double Marchand balun dipole. Rather, it is presented to show how the Marchand baluns may be adjusted to tune the dipole. Parameters Z oc  and Z sc  are related to the widths of open and short circuited stubs, while parameters l oc  and l sc  are related to the lengths of the open and short circuited stubs. Numerous simulations have been performed for various double Marchand balun dipoles. In general it has been noted that as the capacitive loading is increased, range of stub parameter which will produce acceptable match decreases. A bandwidth of three to one with a reflection less than −10 dB is obtainable with reasonable dimensions. A design with a 2.5 to 1 bandwidth extending into both the high and low frequencies regimes was constructed. Designs with bandwidths as great as four to one, entirely in the low frequency regime have been simulated. However, the dimension of such designs may be difficult to realize in practice. 
     Stripline Embodiment 
     Referring back to  FIG. 4 , stripline and microstrip are two common circuit design topologies used in microwave circuit design. All the diagrams of double Marchand balun dipole antennas presented above have featured microstrip designs. This was done to make the diagrams simpler. Stripline, however, may be constructed so that it is symmetrical about the plane of the circuit trace. When applications requiring polarization purity are required, a stripline double Marchand balun dipole design may be preferred. An exploded view of a stripline dipole  160 , comparable to the microstrip dipole shown in  FIGS. 11 and 8 , is shown in  FIG. 13 . Double Marchand balun dipole  160  consists of a first conducting dipole layer  162 , a first dielectric substrate  164 , a conducting feed circuit layer  166 , a second dielectric substrate  168 , and a second conducting dipole layer  170 , all respectively sandwiched together in that order. 
     Feed circuit  166  consists of a tee  172 , feedlines  174  and  176 , and open circuited stubs  178  and  180 . Nominally feedlines  174  and  176  have the same electrical contours although they may follow different contours. The same applies to the open circuited stubs  178  and  180 . 
     The first conducting dipole layer  162  is bifurcated at its midpoint by slot  190  which leads to slot tee  192 . Slot tee  192  feeds short circuited slotline stubs  194  and  196 . Nominally the short circuited stubs  194  and  196  have the same electrical length. Feed lines  174  and  176  cross short circuited stubs  194  and  196 , in opposite directions, at symmetrical distances, which are kept as short as possible, from slot tee  192 . 
     The second conducting dipole  170  is bifurcated at its midpoint by slot  198  which leads to slot tee  200 . Slot tee  200  feeds short circuited slot line stubs  202  and  204 . Nominally dipoles  162  and  170 , and all the features and contours contained within are identical and aligned with each other while being mutually offset through the thickness of dielectric layers  166  and  168 . 
     Conductors  206  and  208  join dipoles  162  and  170  across top edges of dielectric substrate layers  164  and  168 . Electrically dipole layers  162  and  170  and all the features contained within act as one unit. 
     Open circuited stub  178 , and short circuited stubs  194  and  202  constitute a first Marchand balun which corresponds to Marchand balun  106  in  FIG. 9 . Open circuited stub  180 , and short circuited stubs  196  and  204  constitute a second Marchand balun which corresponds to Marchand balun  108  in  FIG. 9 . Slot tees  192  and  200  form an E-plane tee which corresponds to E-plane tee  100  in  FIG. 9 . Finally slots  190  and  198  form a slot line which corresponds to balanced transmission line  98  in  FIG. 9 . 
     Finally substrates layers  164  and  168  extend beyond conducting dipole layers  162  and  170  by to form symmetrical taps  220  and  222 . These preclude electrical contact between conducting dipoles  162  and  168  and corresponding features on adjacent dipoles. 
     The calculated response for the dipole shown in  FIG. 13 , with a unit cell size  224  of 2000 mils, is presented in  FIG. 14 . 
     End and Edge Elements 
     Frequently the end elements in linear arrays and edge elements in planar arrays are not as well matched as interior elements. This is because the elements are designed to accommodate the effects of mutual coupling. The edge and end elements have some of the mutual coupling they have been designed for removed, and as a result function poorly. The simulations discussed earlier used software waveguide simulators to model the effects of mutual coupling. The perfect electric walls are mirrors. The elements, optimized in the presence of mirrors, do not function properly when the mirrors are removed. 
     A common practice in array design is make the end and edge elements into dummy elements. They are match terminated at their ports with resistors, but they are not connected to any beam forming network. Their only purpose is to provide mutual coupling to the next tier of elements in the array. The number of dummy elements to use in an array depends on the level of mutual coupling, the sensitivity of the system to mismatches, and other space and cost considerations. 
     Referring to  FIG. 15A , a double Marchand balun dipole at the end or edge  232  of an array  230  can have its impedance match improved by placing a grounded conducting tab  234  on the outside edge of cell boundary in the plane of the dipole. The configuration shown in  FIG. 15A  may require too much memory for some numerical simulation tools.  FIG. 15B  shows a substitute configuration for designing the end tabs. A double Marchand balun  232  with the relative dimensions discussed above with respect to  FIG. 13 , and with conducting tabs  234  at each end is shown. The height of the tab was chosen to be the same as the height of dipole  232 . The separation between the tab and the dipole was chosen to be the same as the inter-element spacing, and the width of the tab was found empirically with a numerical simulator. 
       FIG. 16  shows the calculated response  240  for the element without conducting end tabs and the calculated response  242  for the element with conducting end tabs. These can be compared to  FIG. 14  which gives the calculated response for same dipole in an infinite planar array environment. 
     It should be noted that  FIG. 15B  depicts a situation worse than what an end or edge element would actually experience. In  FIG. 15B  there are no other elements in the E-plane, but in an array there would be neighboring elements on one end. Thus the response  242  is worse than what would be encountered. The actual responses would be somewhere in between what is shown in  FIG. 14  and  FIG. 16 . As a general rule, −10 dB reflection is the dividing line between matched and unmatched. The curve for conducting tabs in  FIG. 16  is almost acceptable, so in actual array environment, with mutual coupling from one side, the end element  232  in  FIG. 15A  might be acceptable as an active element. 
     This method of end element response enhancement was tested experimentally. An eight element array was constructed with conducting tabs placed adjacent to the end elements. The measured array gain and patterns were consistent with what would be expected for an eight element array, not a six element array, indicating that the end elements were functioning well. 
     Dual Polarized 
     The double Marchand Balun dipole is well suited to radiate two perpendicular linear polarized radiation patterns with coincident phase centers. This is a result of the highly symmetric nature of its construction. Considering the stripline embodiment first because it has perfect symmetry, refer to  FIG. 17A . Here the construction of two perpendicular dipoles, following the method described in  FIG. 13  is shown. The first dipole,  250 , has an upward protruding slot  252 , which begins at the lower edge of the element  254 , and follows element center  256  to some intermediate point  258 . Point  258  is chosen to be sufficiently far from feed line  176  that no disruption of signals propagating on it will occur. Several conducting vias  260 , are added to the construction of  FIG. 13  as follows. The vias pass through dielectric layers  164  and  168  to make electrical contact between conducting dipole surfaces  162  and  170 . The vias  260  are positioned so that their centers lie along element center  256 . The topmost via  262  is positioned as close to E-plane tee,  153 , as construction techniques will permit. The bottommost via  264  is positioned sufficiently far from feed line  176  that no disruption of signals propagating on it will occur. The second dipole  266  follows the construction in  FIG. 13  with the following modification: A downward protruding slot  268  is cut from the bottom edge of E-plane tee  270 , to the same intermediate point  258  previously specified, along element center  256 . The path of feed line  272  is lowered to some intermediate level such that signals propagating on it will not be disrupted by proximity to downward protruding slot  268  or edge  274 . The width of slots  252  and  268  are made slightly larger than the thicknesses of elements  250  and  266 . 
     In  FIG. 17B , the assembly of dipoles  250  and  266  is shown. Dipole  250  passes through slot  268  and dipole  266  passes through slot  252  so that the tops of elements  250  and  266  are flush. Finally, the four seams  280  between the dipoles are soldered with attention to getting the seams to extend as close to E-plane tees  153  and  270  as possible. 
     The procedure for arranging microstrip double balun dipoles in a coincident phase center configuration is detailed in  FIGS. 18A and 18B . Referring to  FIG. 18A , a first microstrip double Marchand balun dipole  290  has a conducting tab  292  added in the region where a second dipole will interleave with it. The tab is made as large as possible while maintaining sufficient distance from balun circuitry so as not to interfere with its performance. In particular, the top edge of the tab  294  is made flush with the bottom edge of the E-plane tee  296  (hidden) on the far side. Several vias  298  between the tab and the dipole conductor are made in the dielectric and plated with conducting material so as to make electrical contact between the tab and the dipole conductor  300  (hidden). The topmost via  302  is made as close to top edge  294  as fabrication technology will allow. The center line of the vias  304  is contiguous with the center of dielectric layer  306  of the second dipole. A notch  308  is cut in the bottom edge of element  290 . The back edge of the slot  310  is positioned so as to provide mechanical clearance for the second dipole circuit board  312 . The front edge  314  and top edge  316  of the slot are positioned so to provide electrical clearance for the feed line on the second dipole when it is joined with the first. 
     A second microstrip double Marchand balun dipole  312  has a downward protruding slot  318 , cut from top edge  320  to a point  322  corresponding to the top edge of slot  316  in the first dipole. The width of slot  318  is just sufficient to provide clearance for the first dipole  290 . At the bottom of slot  318 , there is an enlargement  324  which provides electrical clearance for feed line  326  on the first dipole. The center of the dielectric layer  306  lines up with the center line  304  of the vias on the first dipole. This may be slightly offset from overall center of element  328 . 
       FIG. 18B  shows the assembly  340  of dipoles  290  and  312  into a coincident phase center unit cell. The seam  342  between tab  292  and the dipole conductor of dipole  312  is soldered as is the seam  344  between the dipole conductor of element  290  and element  312 . In both cases special attention is required to make the solder connection extend as close to E-plane tee  296  as possible. 
     It should be noted that the stripline double Marchand balun dipole has both physical and electrical symmetry while the microstrip double Marchand balun has only electrical symmetry. Furthermore, the coincident phase center configuration increases the asymmetry of the microstrip dipole. Numerical simulations have shown that the stripline coincident phase center dipole configuration may have polarization purity in the range of 50 to 60 dB while the microstrip coincident phase center dipole may have polarization purity of 30 to 40 dB. 
     However, for a given dielectric thickness and transmission line impedance wider circuit traces are used with microstrip than with stripline. All the figures presented so for designs that were designed for a two inch cell size which corresponds to a low frequency regime beginning at 3 GHz. If the designs were scaled to 18 GHz, they would be unrealizable in stripline. The thickness of the dipoles would scale down to about 10 mils. Using a low dielectric constant dielectric, Duroid for example, the feed lines of the balun circuit would be about 1.5 mils wide in stripline and about 8 mils wide with microstrip. 
     Isolated Element 
     In simulations of an isolated element, the low frequency response disappears, but the dipole remains well matched in the high frequency regime. Numerous simulations have failed to find any combination of stub dimensions which produce a good match in the low frequency regime. However, combinations of open circuit  94  and  96 , and short circuit,  102  and  104 , stub lengths are easy to find which produce a good match in the high frequency regime. Ultimately the radiation characteristics degrade, limiting the high frequency regime. At the boundary between the low and high frequency regimes, the dipole is a quarter wavelength from the ground plane behind it, which is the optimum condition. At twice that frequency the dipole is one half wavelength above the ground plane behind it which produces a radiation pattern null on axis. 
     Construction Details 
       FIG. 19  shows construction details of double Marchand balun dipole in an array environment. Previously mention was made of the high and low frequency regimes. The division between the two is the frequency at which unit cell size  350  is a half wavelength. Designate that frequency f h  and the corresponding length λ h /2. All of the dipoles presented to this point had a height λ h /4. 
     Generally the low frequency regime is useful for electronic beam steering. Maintaining a perfectly square lattice is only necessary for coincident phase center applications. The most important detail to observe is maintaining electrical continuity across the E-plane tee  352 . 
     Bandwidth Enhancement 
     The upper frequency useful for electronic beam steering is flexible. The smaller the range of scan angles, the higher the cutoff frequency. The frequency f G  at which grating lobes begin to from is given by 
     
       
         
           
             
               f 
               G 
             
             = 
             
               c 
               
                 d 
                 ⁡ 
                 
                   ( 
                   
                     1 
                     + 
                     
                       cos 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       
                         ( 
                         θ 
                         ) 
                       
                     
                   
                   ) 
                 
               
             
           
         
       
     
     where c is the speed of light, d is the element spacing, and θ is the angle of scanning from broadside. Using f G  as upper limit for scanning can increase scanning bandwidth. 
     As noted in reference to  FIG. 11 , bringing the tips of adjacent dipoles closer together improves the low frequency response. In general this causes an increase in reflection at higher frequencies. 
     Increasing the height of the dipoles also improves the low frequency response. In general this also causes an increase in reflection at higher frequencies. 
     Hence extending the bandwidth at the low end results in a compromise. 
     Simulations (see  FIG. 21 ) show that the double Marchand balun is capable of operating over a 4:1 bandwidth if end capacitors similar to those used by Munk in  FIG. 5  are employed and if the height of the dipoles is increased to 3λ h /8. This is shown in  FIG. 20  for a double balun dipole array with a 4200 mil cell size. The distinction with Munk&#39;s design is the that the double Marchand balun dipole has the balun integrated with the dipole so that unbalanced transmission line modes are presented at the ports of the dipole. Munk&#39;s dipoles present balanced transmission line modes at their ports, and this causes a practical difficulties already noted. 
     Obviously many modifications and variations of the present invention are possible in the light of the above teachings. It is therefore to be understood that the scope of the invention should be determined by referring to the following appended claims.