Abstract:
A synchronous rectifier using current driven approach is disclosed which can replace diode rectifier in most of the power converter topologies to enable low rectification loss. The present invention comprises a low loss switch and essentially a transformer with at lease one current sensing winding, windings for current sense energy recovery and one driving winding connected to a hysterisis driver which provides driving signal and power for the synchronous rectifier. A hystersis driver is introduced which can reduce the noise interference to the driving signal, increase the operating frequency range, eliminate the saturation problem of the current sensing transformer and hence provide more flexibility to the transformer design. This synchronous rectifier is self-driven and the driving signal is independent of the input voltage of the converter which enhances its application to wide input range converter. Current sense energy recovery enables power converters to operate at high efficiency and high frequency.

Description:
FIELD OF THE INVENTION 
     This invention relates to the field of power converter, in particular to the field of synchronous rectifier for high efficiency converters. 
     BACKGROUND OF THE INVENTION 
     Power converter designs based on diode rectifiers are limited by the conduction loss of diode rectifiers due to their forward voltage drop, typically 0.7V for silicon diode, during forward conduction. This loss is significant when the rectified output voltage is low and comparable to the forward voltage drop of the diode rectifier. For example, the supply voltage for logic circuits nowadays and microprocessors can be as low as 2.2V or even lower in the future. The output diode rectifiers used in the converters for such applications typically consume one-third of the output power. 
     A known way to improve the rectification efficiency is to replace the diode rectifier by a synchronous rectifier using an active switch with low conduction loss like MOSFET. A synchronous rectifier has a lower forward voltage drop than a diode due to the much lower drop in a transistor. However, as an active switch a synchronous rectifier needs a driving signal in to turn it on at the appropriate times. In addition, the loss and performance of the active switch is sensitive to the driving signal amplitude and waveform. Consequently, the driving method becomes an important issue in synchronous rectifier design. 
     A typical synchronous rectifier makes use of a voltage signal derived from the main transformer windings to drive a MOSFET to ensure that the MOSFET turns on and off in synchronism with the alternating voltage signal on the transformer. However this driving method is not suitable for certain converter topologies. An example is the forward switching regulator using resonant reset. In this case the synchronous rectifier cannot obtain a driving signal during the entire conduction period because the driving voltage collapses with resetting of the main transformer. In the presence of the leakage inductance of the main transformer, no driving signal can be obtained during the commutation period. During this period the body diode instead of the conduction channel of the MOSFETs turns on for current conduction. This increases the losses in the synchronous rectifier especially at high frequency and high current because the forward voltage drop of the body diode is even higher than that of a conventional diode rectifier and with a further increase in the commutation time with the higher output current. Another example of a converter topology that is not driven well by transformer primary/secondary windings is the use of synchronous rectifier in low frequency AC rectification. The slow rising edge of sinusoidal driving voltage, e.g., 50 Hz or 60 Hz main transformer driven by a sinusoidal voltage, cannot efficiently drive the synchronous rectifier into its on state during the conduction period. These limitations impose restrictions on the input voltage range, the choice of topologies of the converter and particular applications. 
     Considerable effort has been expended in tackling the problem of efficiently driving a synchronous rectifier. U.S. Pat. No. 5,179,512, issued to Fisher et al. on Jan. 12, 1993, disclosed a gate drive circuit for synchronous rectifier. However, this gate drive can only work in resonant converters. U.S. Pat. Nos. 5,126,651 and 5,457,624, Kim R. Gauen (issued on Jun. 30, 1992) and Roy A. Hastings (issued on Oct. 10, 1995) respectively disclose drive circuits for synchronous rectifiers. These drive circuits can only be applied to non-isolated buck converter. Similarly, U.S. Pat. No. 5,303,138, issued to Allen F. Rozman on Apr. 12, 1994, disclosed gate drive circuits but without solving the problem of expanding the limited input voltage range. U.S. Pat. No. 5,097,403, issued to David A. Smith on Mar. 17, 1992, disclosed current sense rectifier and electronic circuits to detect current that are only applicable to MOSFET with current sense facility. Notably, U.S. Pat. No. 4,922,404, issued to Ludwig et al. on May 1, 1990, discusses the complexity of using a microprocessor to drive synchronous rectifiers. U.S. Pat. No. 6,134,131, issued to Poon et al. on Oct. 17, 2000, disclosed a current transformer for sensing the current and providing a suitable gate drive for the synchronous rectifier with current sense energy recovery. Although this design is superior in many respects, it is limited by the requirement that the current transformer should not experience saturation due to large operating duty cycle or low operating frequency. Moreover, noise may further interferes with the driving signal. 
     SUMMARY OF THE INVENTION 
     A method and system for improving synchronous rectifier performance with the aid of an additional hystersis driver is disclosed. This driver reduces noise interference with the driving signal, increases the operating frequency range, enhances the driving capabilities even with an otherwise too low a magnetizing inductance to sink the driving current to the gate of the MOSFET. The disclosed method and system overcomes problems due to saturation of the current sensing transformer in addition to producing a low magnetizing inductance resulting in greater flexibility in transformer design. 
     The disclosed method and ssytem encompasses efficient rectification of current in a selected branch of an electronic circuit. It makes use of a low loss MOSFET and with associated circuitry to realize the equivalent of a low loss diode with energy recovered from the current sensing means to ensure high efficiency. 
     In particular, the disclosed embodiments have a low loss active switching device with parallel diode such as a MOSFET, a plurality of windings, two diodes which are connected to a voltage source such as the output voltage or a zener diode. A first winding of the transformer is coupled in series with the diode simulating switching device. A second winding of the transformer is coupled to a hysterisis driver with its output coupled to the control terminal of the switching device. A third and a fourth winding of the transformer each with a series diode are connected to a voltage source. 
     Current flows through the first winding and a series MOSFET. A voltage is induced on the second winding and provides a driving signal for this MOSFET. The second winding is arranged to provide a positive voltage signal to the input of the hysterisis driver so that the MOSFET is driven ON for as long as possible while the current through the first winding flows in the forward direction. 
     A main current flowing through the first winding and the MOSFET produces a voltage on the second winding that, in turn, turns the MOSFET ON. However, this voltage may not be sustained throughout the period during which the current flowing through the primary winding is flowing in the forward direction. This is because magnetizing current increases with time and the voltage collapses when the magnetizing current exceeds the main current in the first winding. Therefore, increasing the time for which the MOSFET is ON improves the efficiency. 
     Use of a hysterisis driver is disclosed as one strategy to turn ON the MOSFET for a longer duration. The hysterisis driver overcomes this limitation because it has preset upper and lower thresholds. It turns on the MOSFET when the voltage induced on the second winding exceeds the upper threshold. Moreover, with the lower threshold set sufficiently low, the turn on signal is maintained as long as the main current remains positive. In other words even after the voltage on the second winding has collapsed the driving signal for turning the MOSFET ON is maintained. This ensures availability of a sufficient driving signal even when the current sensing transformer runs into saturation. Hence, the use of current driven technique results in the synchronous rectifier operating like a low loss active diode with the turning ON and OFF of the active switch independently of the input voltage. 
     The third winding limits the voltage generated and provides energy recovery. Voltage applied to the input of the hysterisis driver as well as the control terminal of the switching device must be limited to avoid damage to the switching device. The third winding of the transformer couples excessive energy to a voltage source and provides voltage clamping. The driving voltage amplitude is controlled by the turn ratio of the second to the third windings and the voltage source. A diode placed in series with the third winding ensures that voltage clamping is effective while the MOSFET is turned ON. This arrangement makes the driving signal independent of input voltage range and waveforms. Excessive energy in the first winding is transferred to the voltage source such as the DC output voltage of the power converter with the recovered current sensing energy becoming part of the output power. 
     The fourth winding provides magnetic reset. A reset mechanism is needed to allow an opposite voltage in the windings in the turn-off period after the transformer is energized during the turn-on period. The fourth winding provides a reset path whereby magnetizing energy stored in the transformer is released through a series connected rectifier to a voltage source. The phase of this fourth winding should be opposite to that of the third winding such that one winding facilitates the turn-on period while the other facilitates the turn-off or reset period. This arrangement allows the magnetizing energy to be recovered and reused. 
     Accordingly, the disclosed method and system provide an improved self-driven synchronous rectifier circuit with current sensing and suitable for wide input voltage and/or frequency ranges. In particular, sufficient driving signal is provided during current commutation along with energy recovery from current sensing. Moreover, the disclosed method and system have application to both isolating and non-isolating converters. 
    
    
     These and other advantages of the present invention will become apparent to those having ordinary skill in the art from the following detailed description of the invention and from the accompanying drawings. 
     BRIEF DESCRIPTION OF DRAWINGS 
     FIG. 1 is a schematic diagram of self-synchronized rectifiers driven by the output voltage of a forward converter. 
     FIG. 2 is a circuit of a basic embodiment of the present invention. 
     FIGS. 3A-3C are timing diagrams of operating voltage and current of the basic embodiment. 
     FIG. 4 is a basic embodiment with less transformer windings and more rectifying diodes. 
     FIG. 5 is a first practical implementation of the present invention. 
     FIG. 6 is a second practical implementation of the present invention. 
     FIG. 7 is a third practical implementation of the present invention. 
     FIG. 8 is a fourth practical implementation of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     The features of the present invention may be better understood by means of the following description of FIG. 1 that shows a typical configuration for driving two synchronous rectifiers on the output stage of a typical forward type converter. The configuration includes a main transformer with its primary winding  9 , secondary-winding  10  and output leakage inductance  8 . The synchronous rectifiers are MOSFETs  1  and  2 , and diodes  3  and  4  are the inherent body diodes of MOSFET  1  and  2  respectively. Inductor  5  and output capacitor  6  put together an output filtering circuit. Resistor  7  represents the equivalent load. 
     Upon application of an alternating voltage to terminals  11  and  12  of primary winding  9 , an alternating voltage is induced in secondary-winding  10 . When the voltage across secondary-winding  10  becomes positive, it drives a current through equivalent leakage inductor  8 . At this time continuous current flows through output inductor  5  and diode  4 . As current through equivalent leakage inductor  8  rises from zero towards the current level of inductor  5 , diode  3  conducts simultaneously with diode  4 . Since the gate terminals of MOSFETs  1  and  2  are connected to the drain terminals of each other, both active switches  1  and  2  are turned off with current flowing through their body diodes in this period. Body diode of MOSFET has a high (with high dissipation) forward voltage of 0.7V that is higher than that of a MOSFET that has been switched ON. When current through diode  3  reaches the current level of inductor  5  diode  4  turns off and MOSFET  1  is allowed to turn on. Only at this time can current flow through the low-loss MOSFET  1  instead of its body diode. 
     When the input alternating voltage goes from positive to negative, the voltage across secondary-winding  10  becomes negative. However, the current flowing through equivalent output leakage inductance  8  cannot drop to zero instantly. Instead there is a period in which simultaneous conduction of MOSFET body diodes  3  and  4  occurs. In this period MOSFETs  1  and  2  are turned off and current flows though the high loss body diodes until current through leakage inductance  8  falls to zero and MOSFET  2  comes into full conduction. 
     The synchronous rectifier so described has three major drawbacks. Firstly simultaneous conduction periods so described reduce converter efficiency. This reduction in efficiency is further aggravated when converter switching frequency and output current increase and/or the simultaneous conduction period becomes a significant portion of the switching period. Secondly, the driving signal for the two MOSFETs is dependent on the voltage and waveform across the secondary-winding of the main transformer. When the supply voltage on the primary side varies over a wide range, the secondary voltage may exceed the gate voltage limit of the MOSFET, or it may be too low to turn the MOSFETs fully on. Thirdly, a special optimum reset mechanism of the main transformer is required to ensure a complete driving signal for the synchronous rectifier or the driving voltage will collapse after the main transformer has been reset and further increase the body diode conduction period and hence the loss. 
     FIG. 2 illustrates an exemplary circuit diagram depicting a MOSFET  100  having parallel diode  105  with its anode connected to the MOSFET source terminal and cathode connected to the MOSFET drain terminal. Although not intended to be a limitation, parallel diode  105 , in general, is the body diode of MOSFET  100 . The circuit diagram further depicts transformer  111  with four windings  101 - 104 . Winding  101  has one end coupled to a terminal  109  and the other end coupled to the source terminal of MOSFET  100 . Winding  102  has one end coupled to the input of hysterisis buffer  112  and the other end to the source terminal of MOSFET  100 . The output of hysterisis buffer  112  is coupled to the gate of MOSFET  100  to provide driving signal. Diode  113  and capacitor  114  form a rectifying circuit to obtain power from winding  102  and provide a DC supply voltage for the hysterisis buffer  112  with voltage approximately equals to the positive amplitude of the voltage across winding  102 . 
     In FIG. 2, winding  103  has one end coupled to the anode of a diode  106 , while the other end couples to the negative terminal of a voltage source  108 . Winding  104  has one end coupled to the anode of a diode  107 , while the other end couples to the negative terminal of voltage source  108 . Diodes  106  and  107  have their cathodes connected together and to the positive terminal of voltage source  108 . Terminal  110  is coupled to the drain terminal of MOSFET  100 . 
     Theory of operation of the first basic embodiment is described. The basic embodiment resembles a diode with anode at terminal  109  and cathode at terminal  110 . When voltage at terminal  109  is higher than that at terminal  110  by a magnitude of the forward voltage drop of diode  105 , current will start to flow from terminal  109  to terminal  110  through winding  101  and body diode  105 . FIG. 3A shows the operating waveforms of I 100  the current flowing through winding  101 , V GS100  the output of the hysterisis buffer  112 , V 102  the input of the hysterisis buffer  112  and I mag111  the magnetizing current of the transformer  111  in this embodiment. As current flows through the current sensing winding  101 , a positive voltage Vg_on will be induced at winding  102 . Winding  102  is arranged so that a positive voltage is induced across the input of hysterisis buffer  112  and source terminal of MOSFET  100 . The upper threshold V H  of hysterisis buffer  112  is set below voltage Vg_on so that the output of the hysterisis buffer equals Vg_on for driving the MOSFET ON and allow current flow through its low resistance channel rather than the body diode  105 . The time interval between current starting to flow through the body diode and the turn on edge of the MOSFET is inversely proportional to product of the magnitude of the operating current and the current gain of the hysterisis buffer, the gate charge required to turn on the MOSFET and the inherent buffer turn on delay. The driving voltage Vg_on is determined by the winding ratio of windings  102  and  103 , the magnitude of the voltage source  108  and coupling coefficient of transformer  111 . Winding  103  is arranged such that current induced in this winding will deliver current into voltage source  108  and the magnitude of this current is determined by the ratio of windings  101  and  103 . Voltage source  108  acts as a voltage clamping facility to clamp the drain source voltage of MOSFET  100 . This mechanism can also recover energy back to voltage source  108 . 
     Turn off operation of the synchronous rectifier is described with reference to FIG.  3 A. When current flowing from terminal  109  to terminal  110  falls to zero, transformer  111  resets itself and generates a negative voltage Vg_off across winding  102 . With V L , the lower threshold of the hysterisis buffer set higher than Vg_off, the output of the hysterisis buffer falls to zero or Vg off, depending on the design of the hysterisis buffer, and drives the MOSFET  100  OFF. The turn off voltage Vg 13  off is determined by the winding ratio of windings  102  and  104 , the magnitude of voltage source  108  and coupling coefficient of transformer  111 . Winding  104  is arranged such that current is delivered to voltage source  108  in the reset process with the magnitude of the current is determined by winding  104  and the magnetic properties of transformer  111 . This charging current actually recovers magnetizing energy stored in transformer  111  and delivers it to voltage source  108 . 
     Voltage source  108  has not been specifically identified but in fact it can be any voltage source with a constant voltage within a converter system. One obvious voltage source is the output of a converter since it allows energy recovered from the current sense winding and energy stored in transformer  111  to be directly utilized by the output load. 
     Losses associated with diodes D 106  and D 107  are low because the current handled by these two diodes are scaled down by the turn ratio of winding  103  to winding  101  and winding  104  to winding  101  respectively. However, the losses can be further reduced by using other low loss switches such as MOSFET with suitable drive. 
     The hysterisis buffer  112  not only reduces the noise problem but also makes the current transformer design flexible. In adverse conditions, such as long duty cycle or high temperature, transformer  111  may be driven to saturation. Driving voltage at winding  102  collapses as a consequence but the normal gate drive signal is not affected in the presence of the hysterisis buffer. 
     FIG. 3B shows the operating waveforms when the transformer is driven beyond saturation. I 100  is the current through winding  101 , V GS100  is the output of the hysterisis buffer  112 , V 102  is the input of the hysterisis buffer  112  and I mag111  is the magnetizing current of the transformer  111 . With transformer  111  driven to saturation, the drive signal across winding  102  or the input to the hysterisis buffer  112  falls to zero. Since the output of the hysterisis buffer  112  changes only when its input falls below the lower threshold V L , the gate drive signal amplitude is maintained if V L  is set to a negative value. When the current flowing through the current sense winding  101  falls sufficiently to reduce the magnetizing current of transformer  111  below the saturation level, the drive signal across  102  becomes negative. The is drive signal triggers the hysterisis buffer  112  to turn off the synchronous rectifier in response to reaching the lower threshold V L . It can be seen that a complete gate drive waveform for MOSFET  100  remains intact. 
     FIG. 3C shows another possible condition of transformer  111 . If the operating duty cycle is long so that the reset period is not enough to reset transformer  111  by the reset voltage of transformer  111 , a high DC component of the magnetizing current sustains. The voltage across winding  102  or the input of the hysterisis  112  will collapse if the magnetizing current is higher than the reflected driving current at this winding and transformer  111  is driven to saturation. Buffer  112  maintains normal gate drive as long as its input does not fall below V L . 
     The operating frequency of the present invention can be as low as AC line frequency range or even lower because the present invention eliminates the need to consider the saturation problem. In other words, the size of transformer can be greatly reduced. 
     Advantageously, no timing circuit or control circuit is needed to generate the necessary synchronous driving signal for MOSFET  100 . 
     FIG. 4 shows another circuit diagram for yet another exemplary embodiment. It differs from FIG. 2 in the usage of four diodes for rectification to a voltage source. The circuit illustrated in FIG. 4 depicts a MOSFET  150  as the main switch having a parallel diode  151  with its anode connected to the MOSFET source terminal and cathode connected to the MOSFET drain terminal. Although not intended to be a limitation, parallel diode  151 , in general, is the body diode of MOSFET  150 . A terminal  157  is coupled the drain terminal of the MOSFET. The circuit diagram further depicts a transformer  155  with three windings  152 - 154 . Winding  153  has one end coupled to a terminal  156  and the other end coupled to the source of MOSFET  150 . Winding  152  has one end coupled to the input of hysterisis buffer  163  and the other end to the source terminal of MOSFET  150 . The output of hysterisis buffer  163  is coupled to the gate of MOSFET  150  to provide driving signal. Winding  154  has one end coupled to the anode of diode  161  and the cathode of diode  159 , while the other end couples to the anode of diode  160  and the cathode of diode  158 . The anodes of diodes  158  and  159  are tied together and coupled to the negative terminal of voltage source  162 . The cathodes of diodes  160  and  161  are tied together and coupled to the positive terminal of voltage source  162 . 
     Operationally, the circuit functions as a diode with anode at terminal  156  and cathode at terminal  157 . When voltage at terminal  156  is higher than that at terminal  157  by a magnitude of the forward voltage drop of diode  151 , current flows from terminal  156  to terminal  157  through winding  153  and body diode  151 . This current flow through the current sensing winding  153  results in a positive voltage Vg being induced at winding  152 . Winding  152  is arranged so that a positive voltage is induced across the input of hysterisis buffer  163  and source terminal of MOSFET  150 . As the voltage induced at the hysterisis buffer  163  input exceeds the upper threshold V H  of hysterisis buffer  163 , the output of the hysterisis buffer becomes sufficiently positive to drive MOSFET  150  ON and shunt current through its low resistance channel rather than body diode  151 . The time interval between current starting to flow through the body diode and the turn on edge of the MOSFET  150  is inversely proportional to product of the magnitude of the operating current and the current gain of the hysterisis buffer, the gate charge required to turn on the MOSFET  150  and the inherent buffer turn on delay. The driving voltage Vg_on is determined by the winding ratio of windings  152  and  154 , the magnitude of the voltage source  162  and coupling coefficient of transformer  155 . Winding  154  delivers current to voltage source  162  and the magnitude of this current is determined by the ratio of windings  154  and  153 . Voltage source  162  acts as a voltage clamping facility to clamp the drain source voltage of MOSFET  150 . This mechanism can also recover energy back to voltage source  162 . 
     Turn OFF operation of the synchronous rectifier is described next. When current flowing from terminal  156  to terminal  157  falls to zero, transformer  155  resets and generate a negative voltage Vg 13  off across winding  152 . With V L , the lower threshold of the hysterisis buffer  163  set higher than Vg off, the output of the hysterisis buffer becomes zero or Vg_off, depending on the design of the hysterisis buffer, and drives the MOSFET  150  OFF. The Vg_off is determined by the winding ratio of  152  and  154 , the magnitude of voltage source  162  and coupling coefficient of transformer  155 . Winding  154  delivers current to voltage source  162  in the reset process and the magnitude of the current is determined by winding  154  and the magnetic properties of transformer  155 . This charging current recovers energy stored in transformer  155  and the gate charge of MOSFET  150  to voltage source  162 . 
     Voltage source  162  can be any voltage source with a constant voltage inside a converter system. One obvious choice is the output of a converter. This allows energy recovered from the current sense winding and the energy store in transformer  162  to be directly utilized by output loads. 
     Losses associated the four diodes D 158 , D 159 , D 160  and D 161  are low because (1) the current handled by these two diodes are scaled down by the turn ratio of winding  154  to winding  153 ; and (2) the losses can further be reduced by replacing these diodes by low loss switches such as MOSFET with suitable drive. 
     As discussed previously, the saturation problem of the transformer  155  does not affect the gate drive and hence the operating frequency can be as low as AC line frequency range or even lower without requiring a large sized core for transformer  155 . In other words, the size of transformer can be reduced greatly for high frequency operation. Furthermore, no timing circuit or control circuit is needed to generate the synchronous driving signal for MOSFET  150 . 
     FIG. 5 shows an embodiment of the invention deployed in an isolated forward converter with half wave rectification. It shows transformer T 201 , the main output transformer of a forward converter that includes primary winding W 201 , secondary-winding W 202  and equivalent leakage inductance L 203 . One terminal of the secondary-winding is coupled to synchronous rectifier unit  220 . Synchronous rectifier unit  220  comprises all components described in the previously described circuit diagrams of FIGS. 2 or  4 . Although not depicted explicitly, synchronous rectifier units  220  and  230  may have the configuration illustrated in FIGS.  2  and/or  4 . A similar synchronous rectifier unit  230  is coupled to another terminal of secondary-winding W 202  and synchronous rectifier unit  220 . Filter inductor L 201  is coupled to synchronous rectifiers  220  and  230 . An output filter capacitor C 201  is coupled to filter L 201 . Output terminals Vo 203  and Vo 204  are coupled to capacitor C 201  that is, in turn, connected to load resistor R 201 . Terminals for connection to a voltage source in the two synchronous rectifier units are connected to output terminals Vo 203  and Vo 204  respectively. 
     The operation during a positive cycle is described next. An AC voltage is applied to primary winding W 201  and a corresponding AC voltage is induced across secondary-winding W 202 . Only half cycle of the AC output voltage will be rectified and filtered to provide DC output. When secondary-winding W 202  transitions into a positive cycle from its negative cycle, current starts to flow through winding N 201  and body diode DM 201 . Current through winding N 201  induces a voltage in winding N 202 . This voltage drives the input of the hysterisis buffer U 201 . Hysterisis buffer U 201  is connected to the gate terminal of MOSFET M 201  to drive M 201  ON. D 203  and C 202  rectify the AC voltage on N 202  to a DC voltage to supply the buffer U 201 . With a continuous current flowing through inductor L 201 , current flowing through the switch M 201  ramps up while current flowing through switch M 202  ramps down correspondingly. The rate of change of current is determined by output leakage inductance L 203  of transformer T 201 . Since both MOSFETs M 201  and M 202  are conducting, the secondary terminal voltage of transformer T 201  is essentially zero, as most of the voltage drops across output leakage inductor L 203 . Nevertheless, both MOSFETs are turned ON by the current through them and kept to minimum voltage drop with minimal dissipation. This solves the problem of simultaneous conduction through body diodes of MOSFETs in the prior art circuit configuration. After current through M 201  has ramped up to the value of current level in inductor L 201 , current flowing through MOSFET M 202  and winding N 205  falls to zero. With no current through winding N 205 , a negative voltage is produced across N 206  that, in turn, drives the hysterisis buffer U 202  to turn MOSFET M 202  OFF. During rest of the positive cycle current flows through synchronous rectifier unit  220  until voltage at winding W 202  changes. 
     The operation of this implementation during a negative cycle is described next. When secondary-winding W 202  exhibits a negative cycle from its positive cycle, voltage applied across the primary winding W 202  is reversed. Current through MOSFET M 201  decreases. However, leakage inductance L 203  of the transformer T 201  keeps its current in the same direction for a finite time. As a result, both MOSFETs will have current flowing while that through M 201  is ramping down and that through M 202  is ramping up. As both switches are turned on, voltage across the transformer secondary terminals is approximately zero. This mechanism keeps the two MOSFETs in ON state and with minimum voltage drops and losses solving the problem of losses due to simultaneous conduction through MOSFET body diodes. This transition period ends with current flowing in M 202  ramping up to the current level of inductor L 201 . Current in M 201  falls to zero and then is turned off. Current continues to flow through M 202  during rest of the negative cycle. 
     When the voltage across the primary winding is zero during one switching cycle, synchronous rectifier unit  203  can still drive MOSFET M 202  on and take advantage of its low loss characteristics. This is because the present invention is current driven. As long as current is continuous through inductor L 201  transistor M 202  will be kept on. This is in contrast to prior art technology that cannot provide proper voltage drive under this condition, as no voltage is induced on secondary-winding and no driving signal can be provided to the MOSFET. 
     This embodiment rectifies positive cycles and produce a steady DC output voltage, it is apparent to those skilled in the art that if the MOSFETs are connected in reverse manner, negative pulse train will be form and hence resulting in negative output voltage. 
     Operation of the described embodiment is independent of input AC voltage on the transformer primary side because it is current driven and not dependent on input voltage. This allows power converter to operate with high efficiency over a wide input voltage range—a significant advantage over prior art technology. 
     FIG. 6 shows another embodiment in the context of an isolated forward converter with center-tapped full-wave rectification. It comprises transformer T 301 , the main output transformer of a forward converter that includes primary winding W 301 , first secondary-winding W 302  and its equivalent leakage inductance L 302 , and second secondary-winding W 303  and its equivalent leakage inductance L 303 . One terminal of first secondary-winding W 302  is coupled to synchronous rectifier unit  320  comprising components described in FIG. 2 (or, alternatively FIG.  4 ). One terminal of second secondary-winding W 303  is coupled to another synchronous rectifier unit  330  that is, in turn, coupled to synchronous rectifier unit  320 . These two synchronous rectifier units are coupled to filter inductor L 301  coupled to filter capacitor C 301 . One terminal of capacitor C 301  is coupled to the center-tapped secondary-windings of transformer T 301 . Output terminal Vo 303  is coupled to capacitor C 301  and inductor L 301 , while another output terminal Vo 304  is coupled to another terminal of capacitor C 301  and the center tap of the secondary-windings. The synchronous rectifiers have MOSFETs M 301  and M 302  as their main switching devices. Terminals for connection to a voltage source in the two synchronous rectifier units are connected to output terminals Vo 303  and Vo 304  respectively. 
     The operation of this embodiment is described herein. An AC voltage is applied to primary winding W 301  and a corresponding AC voltage is induced across secondary-windings W 302  and W 303 . When secondary-winding W 302  exhibits a positive cycle, secondary-winding W 303  exhibits a negative cycle and reverse biases body diode DM 302 . Consequently, there is no current flowing through current sense winding N 305  and MOSFET M 302  is OFF. At the same time body diode DM 301  is forward biased and current flows through current sense winding N 301 . MOSFET M 301  is turned ON with current flowing through this low loss device. Similarly, when secondary-winding W 303  exhibits a positive cycle, secondary-winding W 302  exhibits a positive cycle and reverse biases body diode DM 301 . Similarly, no current flows through current sense winding N 301  and MOSFET M 301  turns OFF. On the other hand, body diode DM 302  is forward biased and current flows through current sense winding N 305  resulting in MOSFET M 302  turning on and current flowing through this low loss device. As a result, both positive and negative cycles are rectified as a positive voltage that is then filtered and a steady DC source produced at the output terminals. 
     Although, the voltage across the transformer primary winding may become zero in a switching cycle, the synchronous rectifier units can still function as low loss switches. Under this condition, the current flowing in inductor L 301  is shared by two paths, one through MOSFET M 301  and secondary-winding W 302 , and another one through MOSFET M 302  and secondary-winding W 303 . Both MOSFETs are turned ON as they are current driven and conduct current in a low loss manner. 
     As noted previously, operation is independent of input AC voltage on the transformer primary side because the design is current driven and not dependent on input voltage. This enables a power converter to operate with high efficiency over a wide input voltage range- a significant advantage. 
     FIG. 7 shows another embodiment in the context of an isolated current doubler type forward converter. It comprises a transformer T 401  that is the main output transformer of a forward converter that includes primary winding W 401 , secondary-winding W 402  and its equivalent leakage inductance L 405 . One terminal of secondary-winding W 402  is coupled to synchronous rectifier unit  420 . This synchronous rectifier unit comprises all components described in the basic embodiment. The coupling point of the transformer secondary is further coupled to inductor L 401 . Another terminal of the secondary-winding W 402  has a symmetrical arrangement. It is coupled to another synchronous rectifier unit  430  that comprises all components described in the basic embodiment. This terminal is further coupled to inductor L 402 . This inductor is coupled to inductor L 401  with output terminal Vo 404 . One terminal of synchronous rectifier  420  attached to winding N 401  is coupled to a terminal of synchronous rectifier  430  attached to winding N 405 . Output terminal Vo 403  is coupled to this node and output capacitor C 401  is coupled to terminals Vo 403  and Vo 404 . These output terminals are further coupled to load resistor R 401 . Terminals for connection to a voltage source in the two synchronous rectifier units are connected to output terminals Vo 403  and Vo 404  respectively. 
     It should be noted that the synchronous rectifier unit may have the configuration depicted in FIG. 2 or  4 . 
     The operation of this embodiment is described next. An AC voltage is applied to primary winding W 401  and a corresponding AC voltage is induced across secondary-winding W 402 . When secondary-winding W 402  exhibits a positive cycle, body diode DM 401  is turned on. Current flowing through winding N 401  and turns on low loss MOSFET M 401 . Current flows through MOSFET M 401  and on to output load resistor R 401 . Since diode DM 402  is reversed biased, no current flows through MOSFET M 402 . The load current is shared by currents in inductors L 401  and L 402 . When secondary-winding W 402  exhibits a negative cycle, body diode DM 402  is turned ON. Current flowing through winding N 405  turns ON low loss MOSFET M 402 . Diode DM 401  is reverse biased and MOSFET M 401  is turned OFF. Note that this circuit arrangement enables power to be delivered to the load attached to the output terminals in both positive and negative cycles with filtering by capacitor C 401  and inductors L 401  and L 402 . The output voltage is positive at terminal Vo 403  and negative at Vo 404 . 
     Although, the voltage across the transformer primary winding may become zero in a switching cycle, the synchronous rectifier units can still function as low loss switches. Since the synchronous rectifier units are current driven as long as sufficient current flow through the switches M 401  or M 402 , they will be turned ON. Their operations are not impaired by the secondary voltage of the transformer dropping to zero or the presence of transformer leakage inductance L 405 . 
     FIG. 8 shows another embodiment in the context of a flyback type converter. Illustrated is a coupled inductor T 501  including primary winding W 501 , secondary-winding W 502  and its equivalent leakage inductance L 502 . One terminal of secondary-winding W 502  is coupled to synchronous rectifier unit  520  that is further coupled to output capacitor C 501 . Output terminals VoSO 3  and Vo 504  are coupled to the positive and negative terminals of capacitor C 501  respectively. These terminals produce a DC output for connection to load R 501 . Negative terminal Vo 504  is coupled to secondary-winding W 502 . The synchronous rectifier unit  520  has its terminals for connection to a voltage source connected to output terminals Vo 503  and Vo 504 . The hysterisis buffer in this embodiment is realized by diodes D 503  and D 504 , transistors Q 501 , Q 502  and Q 503  and resistors R 502 , R 503  and R 504 . Naturally, other designs for hysterisis buffer circuit with suitable driving capability may be used in the synchronous rectifier unit to enhance gate drive signal for its MOSFET. It should be noted that, among other designs, the synchronous rectifier unit may have the configurations shown in FIG. 2 or  4 . 
     The operation of this embodiment is described next. An AC voltage is applied to primary winding W 501  and a corresponding AC voltage is induced across secondary-winding W 502 . Windings W 501  and W 502  are arranged so that they produced voltage of opposite phase. When secondary-winding W 502  exhibits a positive cycle, body diode DM 501  is turned on. Current flows through winding N 501  and induces a positive voltage across N 502 . When this induced positive voltage goes higher than the forward drop of D 503  (˜0.6V) plus the base to emitter forward bias voltage of Q 502  (˜0.6V), Q 502  will be driven on to turn on a low loss MOSFET M 501  through R 504 . This means the upper threshold of this hysterisis buffer is approximately equal to 1.2V. Current then flows through low loss channel of MOSFET M 501  and delivers current to output load resistor R 501 . When secondary-winding W 502  exhibits a negative cycle, current flowing in winding N 501  during the positive cycle will fall with falling rate proportional to the negative voltage across W 502  and inversely proportional to the leakage inductance L 502 . When this current falls to zero, a negative voltage will be induced across N 502  because of the stored magnetizing energy of T 502  during the positive cycle. When this negative voltage falls to a negative value (˜−0.6V) so that the base to emitter junction of Q 501  is forward biased, Q 501  will then be turned on and hence turn on the transistor Q 503  to discharge the gate voltage of M 501 . This means the lower threshold of this implemented hysterisis buffer is approximately equal to −0.6V. M 501  will then be turned off to cease current flow in opposite direction to that in a positive cycle. Similar to operations in other embodiments the synchronous rectifier provides energy recovery for high efficiency operations. 
     The present invention was subjected to experimental evaluation in a forward converter based design. Two experiments were carried out. In one experiment the secondary section of the forward converter comprised of Schottky diode of type MBR1645, which is a 16A, 45V device. In another experiment the secondary section comprised of the present invention as synchronous rectification units. The switching devices in the synchronous rectification units are MOSFET of type SGS60NE03L-10 having a turn on resistance of 10 milli-ohm. The converter operates under the same condition with a load current of 4 A. In both cases the temperature rise of the devices was recorded. The temperature rise for the Schottky diode was 27 degree C. whereas the temperature rise for the MOSFET was only 6 degree C. These two types of devices have the same package type TO220. This experiment verified the effectiveness of the present invention in reducing losses and increasing the efficiency. 
     While the invention has been described in connection with what is presently considered to be the most practical and preferred embodiments, it is to be understood that the present invention is not limited to the disclosed embodiments and is expressly intended to cover various modifications and equivalent arrangements included within the scope of the appended claims. 
     Thus, it will be appreciated that the various features described herein may be used singly or in any combination thereof. Thus, the present invention is not limited to only the embodiments specifically described herein. While the foregoing description and drawings represent an embodiment of the present invention, it will be understood that various additions, modifications, and substitutions may be made therein without departing from the spirit and scope of the present invention as defined in the accompanying claims. In particular, it will be clear to those skilled in the art that the present invention may be embodied in other specific forms, structures, and arrangements, and with other elements, and components, without departing from the spirit or essential characteristics thereof. One skilled in the art will appreciate that the invention may be used with many modifications of structure, arrangement, and components and otherwise, used in the practice of the invention, which are particularly adapted to specific environments and operative requirements without departing from the principles of the present invention. The presently disclosed embodiment is therefore to be considered in all respects as illustrative and not restrictive, the scope of the invention being indicated by the appended claims, and not limited to the foregoing description.