Abstract:
The present invention relates to a transmitter and power supply which suffers little EMI deviation and provides adequate margin even in volume production, and which can cut the unit price of transformers and economize on the costs of EMI filters, by cancelling out and eliminating the conducted noise or the conducted noise and radiated noise caused by capacitive coupling between the windings of a transformer, the transformer being of an uncomplicated construction which is good in terms of production.

Description:
This application is a U.S. national stage filing of International Application No. PCT/KR2011/008433, filed Nov. 7, 2011, which claims priority to Korean Patent Application Nos. KR 10-2010-0111007 filed on Nov. 9, 2010, KR 10-2011-0000053 filed on Jan. 3, 2011, KR 10-2011-0000051 filed on Jan. 3, 2011, KR 10-2011-0011967 filed on Feb. 10, 2011, KR 10-2011-0030776 filed on Apr. 4, 2011, KR 10-2011-0036294 filed on Apr. 19, 2011, KR 10-2011-0073383 filed on Jul. 25, 2011, KR 10-2011-0073959 filed on Jul. 26, 2011, KR 10-2011-0078198 filed on Aug. 5, 2011, KR 10-2011-0078197 filed on Aug. 5, 2011, KR 10-2011-0084619 filed on Aug. 24, 2011, KR 10-2011-0084620 filed on Aug. 24, 2011, all of which are incorporated herein by reference in its entirety. 
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to a transformer having a simple structure with a high productivity, and more particularly, to a transformer and a power supply for cancelling out conducted noise or conducted noise and radiated noise due to a capacitive coupling between transformer windings to provide a small EMI deviation and a sufficient margin even during mass production, thereby reducing the unit cost of the transformer, and reducing the cost of the EMI filter. 
     2. Description of the Related Art 
     So far, there has been a magnetic energy-transfer element or power supply configured to reduce a displacement current flowing to the electrical ground from the power supply using a cancellation effect due to magnetic energy-transfer element windings. However, five or six strands of thin wire should be stretched and wound to fill one winding layer with no gap using a small number of turns for cancellation. Accordingly, it has may disadvantages in that the winding operation is difficult and the productivity of the transformer is low, thus increasing the unit cost, and in case of a transformer with a low profile form factor, several strands cannot be connected to a pin, and the like. Furthermore, modified shaped methods have been used to get out of the restriction of the height of a transformer having a low profile form factor. In this case, it has a large deviation in the effect of reducing a displacement current flowing to the electrical ground from the power supply, thus causing a disadvantage in that it is difficult to satisfy EMI standards. 
     The prior art will be described below in brief. 
       FIG. 1  is a view illustrating a principle in which the transformer  13 , input line  16  and output line  17  are coupled by a distributed capacitance within the transformer in a typical flyback converter to generate a displacement current to the electrical ground. Hereinafter, a black dot shown for each winding of the transformer indicates the start or end of the winding. 
     Referring to  FIG. 1 , an AC input voltage is rectified and smoothened by the capacitor  11 . The switching element  12  is switched in response to the feedback of the output voltage to create the storage and transferring of energy in the input winding  131  of the transformer  13 , and the output rectifier  14  and capacitor  15  rectifies the voltage of the output winding  133  to supply power to a load. 
     Typically, the varying speed of voltage at the connection point between an end of the input winding  131  of the transformer  13  and the switching element  12  is very fast when the switching element  12  is turned on or off, and the potential variation of maximum 500-600 volts occurs. The potential variation is transferred to the output winding  133  through the path of a distributed capacitance (Cps) between the input winding  131  and the output winding  133  or the path of a distributed capacitance (Cpc) between the input winding  131  and the transformer core and a distributed capacitance (Csc) between the transformer core and the output winding  133 , thus allowing the output line  17  to have a noise potential. The potential variation allows the input line  16  to have a noise potential through a distributed capacitance (Cpi) between the input winding  131  and the input line  16 . Furthermore, the potential variation allows the transformer core to have a noise potential through a distributed capacitance (Cpc) between the input winding  131  and the transformer core  136 . Those noise potentials allow a current to flow through a distributed capacitance (Cig) between the input line  16  and the ground, a distributed capacitance (Cog) between the output line  17  and the ground, and a distributed capacitance (Ccg) between the transformer core and the ground, thereby generating common mode noise, and thus the noise current should be managed to be less than a level specified by the regulations. 
       FIG. 2  is a principle view for cancelling a capacitive coupling of the output winding by a potential of the input winding in the related art. 
     Referring to  FIG. 2 , the input winding  131  generates a capacitive coupling current through a distributed capacitance of the surface facing the output winding  133  by generating an electric field in the direction of facing the output winding  133 , and generates a capacitive coupling current through a distributed capacitance between the input winding  131  and the transformer core  136  and a distributed capacitance between the transformer core  136  and the output winding  133  by generating an electric field in the direction opposite to that of facing the output winding  133 . 
     Referring to  FIG. 2 , a capacitive coupling current between the input winding  131  and the output winding  133  should be maintained low to maintain a displacement current flowing to the electrical ground through the output line. In  FIG. 2 , to this end, an electric field generated in the direction of facing the output winding  133  from the input winding  131  is shielded by the cancellation winding  132 , and an electric field generated in the direction opposite to that of facing the output winding  133  from the input winding  131  is shielded by the shield winding  134 . 
     Furthermore, a capacitive coupling generated in spite of the shielding is removed by the shield winding  134  that forms an electric field using a potential having a polarity opposite to that of the input winding  131 . Furthermore, the cancellation winding  132  generates a capacitive coupling having a reversed polarity between the cancellation winding  132  and the output winding  133 , thereby cancelling out a capacitive coupling between the input winding  131  and the output winding  133  generated in spite of the shielding. 
     In order to generate a current having a reverse polarity for cancelling out a capacitive coupling generated from the input winding  131  having a high potential variation to the output winding  133  having a low potential variation with the same polarity, the cancellation winding  132  should have a potential variation lower than that of the output winding  133 , and to this end, the number of turns (T: turn) of the cancellation winding  132  is less than that of the output winding  133 . 
     For example, a transformer having a winding width of 8 mm, which is widely used for a mobile phone charger power supply with the input of a commercial voltage of 220 V and the output of 5 V is taken as an example. When the number of turns of the output winding  133  is 8T (T: turn), the number of turns of the cancellation winding  132  is 6T to 7T to cancel out the coupling while shielding the input winding  131  from being capacitively coupled to the output winding  133 . In order to completely surround the winding width of 8 mm with 7T, six strands of thin wire having a diameter of 0.18 mm should be uniformly stretched and wound in parallel with no gap, and thus the winding work may be difficult, thereby reducing the productivity and increasing the cost. 
       FIG. 3  illustrates an example of the transformer of  FIG. 2 , and  FIG. 4  is an example further including three strands of bias winding  135  for pulling out an auxiliary power of about 10 V from the transformer of  FIG. 3 . Total nine strands should be connected to a common grounding terminal ( 5   a  and  7   a ) to which three strands of bias winding  135  and six strands of cancellation winding  132  should be connected, but such a method cannot be used for a small-sized product in which the height of soldered components is restricted. 
       FIG. 5  illustrates the structure of a modified transformer for enhancing the productivity of a winding. It has a structure in which the bias winding  135  having a number of turns far greater than that for cancellation is located between the input winding  131  and the output winding  133 , and one strand of cancellation winding  137  capacitively coupled to part of the output winding  133  to cancel out a capacitive coupling generated between the input winding  131  and the output winding  133  and between the bias winding  135  and the output winding  133  is added. However, a barrier tape  138  for holding the location of the cancellation winding  137  has a large width deviation, and the physical location of the cancellation winding  137  is varied, and thus a large deviation occurs at a coupling between the cancellation winding  137  and the output winding  133 . The deviation has a disadvantage in that EMI is generated to a large extent according to the product. 
       FIG. 6  is an example having a sandwich winding structure in the related art, in which it is divided into a first input winding  131   a  having a small potential variation width and a second input winding  131   b  having a large potential variation width of the input winding to surround both the winding surfaces of the output winding  133  in a sandwich shape. The first shield winding  132   a  and second shield winding  132   b  are located between the first input winding  131   a  and the output winding  133  and between the second input winding  131   b  and the output winding  133 , respectively, to shield a capacitive coupling between the first input winding  131   a  and the output winding  133  and between the second input winding  131   b  and the output winding  133 . However, even though a capacitive coupling between the second input winding  131   b  and the output winding  133  having a large potential variation width is shielded, it generates a coupling current far greater than the coupling current occurring between a winding layer having the lowest potential variation width among the winding layers of the input winding  131  in  FIG. 3  and the output winding  133 . Furthermore, a high spike voltage inherent in the second input winding  131   b  having a large potential variation width forms another noise on the second shield winding  132   b . Accordingly, large conducted noise and radiated noise may be generated, thus requiring measures for noise reduction such as reinforcing line filters, using high frequency filters, and the like. 
     According to the related art, six strands should be wound in parallel for a wire, and thus the automation is difficult and the productivity is reduced, and soldering a large number of wires to terminals does not satisfy the height restriction in small-sized products, and the shielding deviation is high when shielding with bias windings and balanced windings to reduce the number of strands of wire, thereby deteriorating the EMI margin management. Furthermore, large conducted noise and large radiated noise may be generated in a sandwich winding structure, thus having disadvantages of requiring measures for noise reduction such as reinforcing line filters, using high frequency filters, and the like. The present invention is contrived to solve all the foregoing disadvantages in the related art. 
     SUMMARY OF THE INVENTION 
     The present invention may be applicable to non-insulation type buck converters, buck-boost converters and boost converters, and insulation type forward converter and flyback converter, but the description according to the embodiments will be mainly described with respect to the flyback converter. 
     In order to accomplish the foregoing object, there is provided a magnetic energy-transfer element used for a switching type power supply comprising a first voltage input terminal, a second voltage input terminal, a switching element, a magnetic energy-transfer element, an output rectifier, and an output line, and the magnetic energy-transfer element may include a core of the magnetic energy-to transfer element; an input winding wound around the core of the magnetic energy-transfer element, wherein the flow of current and the transfer of magnetic energy are switched by the switching operation of the switching element; an output winding wound to face one side surface of the input winding and magnetically coupled to the input winding to take out energy and supply it to the load, wherein the polarity of the potential variation of a terminal connected to the output rectifier is opposite to that of the potential variation at a connecting point between an end of the input winding and an end of the switching element; and a cancellation winding configured to shield a capacitive coupling through the distributed capacitance of a surface facing each other between the input winding and the output winding, and to generate a capacitive coupling to the output winding so as to cancel out and reduce the sum of capacitive couplings generated from windings other than the output winding and the core of the magnetic energy-transfer element to the output winding, wherein the number of turns of the cancellation winding wound per unit area of one winding layer for reducing the sum of capacitive couplings generated to the output winding is greater than that of the output winding wound per unit area of one winding layer. 
     Furthermore, in order to accomplish the foregoing object, there is provided a magnetic energy-transfer element used for a switching type power supply comprising a first voltage input terminal, a second voltage input terminal, a switching element, a magnetic energy-transfer element, an output rectifier, and an output line, and the magnetic energy-transfer element may include a core of the magnetic energy-transfer element; a first input winding wound around the core of the magnetic energy-transfer element, and connected between the first voltage input terminal and one side terminal of the switching element, wherein the flow of current and the transfer of magnetic energy are switched by the switching operation of the switching element; and a second input winding wound around the core of the magnetic energy-transfer element, and connected between the second voltage input terminal and the other side terminal of the switching element, wherein the flow of current and the transfer of magnetic energy are switched by is the switching operation of the switching element, wherein an effect exerted to the outside due to a potential variation and generated noise of the first input winding by the switching operation of the switching element and an effect exerted to the outside due to a potential variation and generated noise of the second input winding by the switching operation of the switching element are cancelled out due to their opposite polarities. 
     Furthermore, in order to accomplish the foregoing object, there is provided a magnetic energy-transfer element used for a switching type power supply comprising a first voltage input terminal, a second voltage input terminal, a switching element, a magnetic energy-transfer element, an output rectifier, and an output line, and the magnetic energy-transfer element may include a core of the magnetic energy-transfer element; a first input winding wound around the core of the magnetic energy-transfer element, and connected between the first voltage input terminal and one side terminal of the switching element, wherein the flow of current and the transfer of magnetic energy are switched by the switching operation of the switching element; a second input winding wound around the core of the magnetic energy-transfer element, and connected between the second voltage input terminal and the other side terminal of the switching element, wherein the flow of current and the transfer of magnetic energy are switched by the switching operation of the switching element; and an output winding magnetically coupled to the first input winding and the second input winding to take out energy, wherein an effect exerted to the outside due to a potential variation and generated noise of the first input winding by the switching operation of the switching element and an effect exerted to the outside due to a potential variation and generated noise of the second input winding by the switching operation of the switching element are cancelled out each other due to their opposite polarities. 
     Furthermore, there are provided a buck converter, a buck-booster converter, a boost converter, a flyback converter, and a forward converter including the foregoing magnetic energy-transfer element according to the present invention. 
     Furthermore, there is provided a manufactured article including the foregoing power supply according to the present invention. 
     Hereinafter, a transformer and a power supply having a structure of cancelling out noise according to the present invention will be described in detail with reference to the accompanying drawings. 
     According to the present invention, a capacitive coupling between the input winding and the output winding of a transformer may be cancelled out, thereby the number of turns of the cancellation winding for reducing noise potential of the output line is further increased, a winding operation with a further smaller number of strands of wire is allowed, facilitating the automation of the winding operation due to simplified winding structure of the transformer, enhancing the productivity, reducing the cost of the transformer, enhancing the deviation of EMI due to a small deviation of cancellation, and easily coping with the height restriction of a soldering portion on a terminal for small-sized products having a component height restriction. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The accompanying drawings, which are included to provide a further understanding of the invention and are incorporated in and constitute a part of this specification, illustrate embodiments of the invention and together with the description serve to explain the principles of the invention. 
       In the drawings: 
         FIG. 1  is a generation diagram illustrating a displacement current flowing to the ground by a distributed capacitance within a transformer in a flyback converter according to the related art; 
         FIG. 2  is a principle diagram of cancellation in the related art; 
         FIGS. 3 through 5  are embodiments illustrating the structure of a transformer in the related art; 
         FIG. 6  is an embodiment to which the related art is applied to a sandwich structure; 
         FIG. 7  is a diagram illustrating Principle  1  of cancellation of capacitive coupling of a transformer according to the present invention; 
         FIG. 8  is an embodiment illustrating a transformer configured according to Principle  1  in  FIG. 7 ; 
         FIG. 9  is a configuration diagram illustrating a flyback converter to which the transformer of  FIG. 8  is applied; 
         FIG. 10  is another embodiment illustrating a transformer configured according to Principle  1  in  FIG. 7 ; 
         FIG. 11  is a configuration diagram illustrating a flyback converter to which the transformer of  FIG. 10  is applied; 
         FIG. 12  is a diagram illustrating Principle  2  of cancellation of capacitive coupling of a transformer according to the present invention; 
         FIG. 13  is an embodiment illustrating a transformer configured according to Principle  2  in  FIG. 12 ; 
         FIG. 14  is a configuration diagram illustrating a flyback converter to which the transformer of  FIG. 13  is applied; 
         FIG. 15  is another embodiment illustrating a transformer configured according to Principle  2  in  FIG. 12 ; 
         FIG. 16  is a configuration diagram illustrating a flyback converter to which the transformer of  FIG. 15  is applied; 
         FIG. 17  is a diagram illustrating Principle  3  of cancellation of capacitive coupling of a transformer according to the present invention; 
         FIG. 18  is an embodiment illustrating a transformer configured according to Principle  3  in  FIG. 17 ; 
         FIG. 19  is a configuration diagram illustrating a flyback converter to which the transformer of  FIG. 18  is applied; 
         FIG. 20  is another embodiment illustrating a transformer configured according to Principle  3  in  FIG. 17 ; 
         FIG. 21  is a configuration diagram illustrating a flyback converter to which the transformer of  FIG. 20  is applied; 
         FIG. 22  is still another embodiment illustrating a transformer configured according to Principle  3  in  FIG. 17 ; 
         FIG. 23  is a configuration diagram illustrating a flyback converter to which the transformer of  FIG. 22  is applied; 
         FIG. 24  is yet still another embodiment illustrating a transformer configured according to Principle  3  in  FIG. 17 ; 
         FIG. 25  is a configuration diagram illustrating a flyback converter to which the transformer of  FIG. 24  is applied; 
         FIG. 26  is still yet another embodiment illustrating a transformer configured according to Principle  3  in  FIG. 17 ; 
         FIG. 27  is a configuration diagram illustrating a flyback converter to which the transformer of  FIG. 26  is applied; 
         FIGS. 28 through 30  are embodiments for suppressing the generation of high frequency noise in a flyback converter to which the transformer of Principle  3  in  FIG. 17  is applied; 
         FIG. 31  is an embodiment illustrating a transformer having a sandwich winding structure according to the present invention; 
         FIG. 32  is a configuration diagram illustrating a flyback converter to which the transformer of  FIG. 31  is applied; 
         FIGS. 33 through 35  are other embodiments illustrating a transformer having a sandwich winding structure according to the present invention; and 
         FIGS. 36 and 37  are configuration diagrams illustrating a forward converter according to the present invention. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     [First Embodiment] 
       FIG. 7  is Principle  1  which is an embodiment of a transformer  19   a  for shielding and cancelling out a capacitive coupling between the input winding and the output winding having a single directional potential variation. 
     According to Principle  7  of  FIG. 7 , the transformer  19   a  is configured with an input winding  191   a , an output winding  193 , and a cancellation winding  192   a  wound around the winding section of a transformer core  196 . 
       FIG. 8  is an embodiment illustrating the transformer  19   a  configured according to Principle  1  in  FIG. 7 , and  FIG. 9  is a configuration diagram illustrating a flyback converter to which the transformer  19   a  of  FIG. 8  is applied. 
     The input winding  191   a  of  FIG. 7  has a potential variation due to a current flow interruption by the switching operation of the switching element  12  illustrated in  FIG. 9 , and the potential variation of a terminal connected to the output rectifier among the terminals of the output winding  193  has an opposite polarity to that of the input winding  191   a.    
     At every instant when the potential of the input winding  191   a  varies according to the switching operation of the switching element  12 , the output winding  193  is capacitively coupled due to electric fields variation caused by the potential variation of the input winding  191   a.    
     In the transformer  19   a  of  FIG. 7 , a capacitive coupling due to a potential variation between the input winding  191   a  and the output winding  193  may be divided into a coupling due to an electric field generated in the direction of the input winding  191   a  facing the output winding  193  and a coupling due to an electric field generated in the opposite direction to the direction of the input winding  191   a  facing the output winding  193 . The transformer core  196  is capacitively coupled by an electric field generated in the opposite direction to the direction of the input winding  191   a  facing the output winding  193 , and the core  196  is capacitively coupled to the output winding  193  again through the magnetic path of the core. 
     In order to cancel out and remove the sum of capacitive couplings generated from windings other than the cancellation winding  192   a  and the transformer core  196  to the output winding  193 , the cancellation winding  192   a  allows a potential difference between the cancellation winding  192   a  and the output winding  193  to generate a capacitive coupling. 
     In other words, the cancellation winding  192   a  allows the size of a capacitive coupling generated from the cancellation winding  192   a  to the output winding  193  due to a potential difference from the output winding  193  to be equal to the sum of capacitive couplings with the opposite polarity generated from windings other than the cancellation winding  192   a  and the transformer core  196  to the output winding  193 , thus cancelling out and removing all capacitive couplings generated to the output winding  193 . 
     The cancellation winding  192   a  should have a potential variation with the opposite polarity greater than that of the output winding  193  to generate a capacitive coupling with the opposite polarity to the capacitive coupling by a potential difference between the output winding  193  having a potential variation with the opposite polarity to that of the potential variation of the input winding  191   a  and the input winding  191   a . Accordingly, the number of turns of the cancellation winding  192   a  for cancellation is greater than that of the output winding  193 . 
     Furthermore, the cancellation winding  192   a  is wound to fill one winding layer between the input winding  191   a  and the output winding  193  with no gap to shield a capacitive coupling due to an electric field generated in the direction of the input winding  191   a  facing the output winding  193 , thus generating a very small capacitive coupling. The cancellation winding  192   a  is wound to fill one winding layer between the input winding  191   a  and the output winding  193  with no gap for shielding. 
     When an electric field generated in the opposite direction to the direction of the input winding  191   a  facing the output winding  193  is not capacitively coupled at all to the output winding  193 , the number of turns of the cancellation winding  192   a  is set to be greater than that of the output winding  193  by 1T-2T to cancel out a small amount of capacitive coupling due to an electric field generated in the direction of the input winding  191   a  facing the output winding  193  and shielded. 
     However, according to the present invention, an amount of capacitive coupling due to an electric field generated in the opposite direction to the direction of the input winding  191   a  facing the output winding  193  is set to be greater than that of an electric field generated in the direction of the input winding  191   a  facing the output winding  193  and shielded, and thus the number of turns of the cancellation winding  192   a  required for cancellation is adjusted to the number of turns with good productivity. 
     As increasing the amount of capacitive coupling due to an electric field generated in the opposite direction to the direction of the input winding  191   a  facing the output winding  193 , a difference between the number of turns of the cancellation winding  192   a  for cancellation and the number of turns of the output winding  193  may be further increased, but if the difference is too large, then conducted noise will increase. 
     The transformer  19   a , which is an embodiment of the present invention according to Principle  1  in  FIG. 7 , will be described below. 
     In the transformer  19   a  of  FIG. 8 , the input winding  191   a , the cancellation winding  192   a  and the output winding  193  are sequentially wound around the transformer core  196 . A winding layer closest to the output winding  193  among the winding layers of the input winding  191   a  is a winding layer having the lowest potential variation among the winding layers of the input winding  191   a . The strength of an electric field generated in the direction of the input winding  191   a  facing the output winding  193  is mainly influenced by the potential of a winding layer having the lowest potential variation among the winding layers of the input winding  191   a . A winding layer located at an opposite end in the direction of the input winding  191   a  facing the output winding  193  among the winding layers of the input winding  191   a  is a winding layer having the highest potential variation among the winding layers of the input winding  191   a . The strength of an electric field generated in the opposite direction to the direction of the input winding  191   a  facing the output winding  193  is mainly influenced by the potential of a winding layer having the highest potential variation among the winding layers of the input winding  191   a.    
       FIG. 9  is an example of a flyback converter to which the transformer  19   a  is applied. 
     In  FIG. 9 , the capacitor  11 , switching element  12 , input line  16  and output line  17  correspond to the elements in  FIG. 1 , respectively. 
     The output rectifier  14   a  of  FIG. 9  rectifies a negative voltage since the output winding  193  of the transformer  19   a  has a potential variation with the opposite polarity to that of the input winding  191   a , and the polarities of the voltage of the capacitor  15  and the output voltage are opposite to those of  FIG. 1 . 
     The present invention described above with reference to  FIGS. 7 through 9  will be summarized again below. 
     The transformer  19   a  of the present invention according to Principle  1  in  FIG. 7  may include a core  196  of the magnetic energy-transfer element; an input winding  191   a  wound around the core  196  of the magnetic energy-transfer element, wherein the flow of current and the transfer of magnetic energy are switched by the switching operation of the switching element  12 ; an output winding  193  wound to face one side surface of the input winding  191   a  and magnetically coupled to the input winding  191   a  to take out energy and supply it to the load, wherein the polarity of the potential variation of a terminal connected to the output rectifier  14   a  is opposite to that of the potential variation at a connecting point between an end of the input winding  191   a  and an end of the switching element  12 ; and a cancellation winding  192   a  located between the input winding  191   a  and the output winding  193  to shield a capacitive coupling through the distributed capacitance of a surface facing each other between the input winding  191   a  and the output winding  193 , and to generate a capacitive coupling to the output winding  193  so as to cancel out and reduce the sum of capacitive couplings generated from windings other than the output winding  193  and the core  196  of the magnetic energy-transfer element to the output winding  193 , wherein the number of turns of the cancellation winding  192   a  wound per unit area of one winding layer for reducing the sum of capacitive couplings generated to the output winding  193  is greater than that of the output winding  193  wound per unit area of one winding layer. 
     Furthermore, in the transformer  19   a  of the present invention according to Principle  1  in  FIG. 7 , an electric field generated from a winding surface of the winding located at an opposite end in the direction of the input winding  191   a  facing the output winding  193  in opposite direction to the direction of the input winding  191   a  facing the output winding  193  is capacitively coupled to the output winding  193  through the core  196  of the magnetic energy-transfer element. 
     Furthermore, in the transformer  19   a  of the present invention according to Principle  1  in  FIG. 7 , as increasing an amount of capacitive coupling generated to the output winding  193  by an electric field formed in the opposition direction to the direction of the input winding  191   a  facing the output winding  193  from the winding surface of a winding layer located at an opposite end of the direction facing the output winding  193  among the winding layers of the input winding  191   a , the number of turns of the cancellation winding  192   a  wound per unit area of one winding layer is further greater than that of the output winding  193  wound per unit area of one winding layer. 
       FIG. 10  illustrates the transformer  19   b  which is another embodiment of the present invention according to Principle  1  in  FIG. 7 . 
     In the transformer  19   a  of  FIG. 8  as described above, a layer having the highest potential variation among the winding layers of the input winding  191   a  is located at an end in the opposite direction to the direction facing the output winding  193  and generates an electric field in the opposite direction to the direction facing the output winding  193 . When an AC 220V input is rectified and used, the potential variation width of a layer having the highest potential variation is too high, approximately 500V, and thus an electric field generated in the opposite direction to the direction facing the output winding  193  may be too large, and a coupling to the output winding  193  may be excessively generated. On the contrary, when a layer having a high potential variation among the winding layers of the input winding  191   a  is closest to the output winding  193  and a layer having the lowest potential variation is located at an end in the opposite direction to the direction facing the output winding  193 , ringing at a high spike voltage contained in the input winding  191   a  may be transferred to the cancellation winding  192   a  through a distributed capacitance, thus causing a problem of generating a cancellation error. 
       FIG. 10  is to reduce the problem of the transformer  19   a  of  FIG. 8  as described above. 
     The transformer  19   b  locates a winding layer  191   b - c  having the highest potential variation among the winding layers of the input winding  191   b  between a winding layer  191   b - a  having the lowest potential variation and a winding layer  191   b - b  having a middle potential variation. The structure may prevent ringing at a high spike voltage contained in the input winding  191   a  of the transformer  19   a  in  FIG. 8  from affecting on other windings through a distributed capacitance. In the transformer  19   b , the winding layer  191   b - a  having the lowest potential variation among the winding layers of the input winding  191   b  is located at an end in the opposite direction to the direction facing the output winding  193  to adjust the strength of an electric field generated in the opposite direction to the direction facing the output winding  193 . 
     When the number of turns is set by differently configuring the thickness or number of strands of wire of the winding layer  191   b - a  located at an end in the opposite direction to the direction facing the output winding  193  from other winding layers  191   b - b  or  191   b - c , the strength of an electric field generated in the opposite direction to the direction facing the output winding  193  by the potential of the winding layer  191   b - a  may be set differently, thereby allowing the number of turns of the cancellation winding  192   b  required for cancellation to be set to desired value. 
     For the transformer  19   b  of  FIG. 10 , the location arrangement of each winding layer  191   b - a  to  191   b - c  in the input winding  191   b  may vary according to the potential variation width of the input winding  191   b.    
     In other words, when the potential variation width of the input winding  191   b  is low, the winding layer  191   b - c  having the highest potential variation among the winding layers of the input winding  191   b  may be located at an end in the opposite direction to the direction facing the output winding  193 . When the potential variation width of the input winding  191   b  is greater than that value, the winding layer  191   b - b  having a middle potential variation among the winding layers of the input winding  191   b  may be located at an end in the opposite direction to the direction facing the output winding  193 , and the winding layer  191   b - a  having the lowest potential variation may be located when it is very high. 
       FIG. 11  is a configuration diagram illustrating a flyback transformer to which the transformer  19   b  is applied, and the elements other than the transformer  19   b  correspond to those of  FIG. 9 . 
     The present invention described above with reference to  FIGS. 10 and 11  will be summarized again below. 
     In the transformer  19   b  of the present invention according to Principle  1  in  FIG. 7 , the location arrangement of the winding layer  191   b - a  having the lowest potential variation, the winding layer  191   b - c  having the highest potential variation and winding layer  191   b - b  having a middle potential variation is selected from the winding layers of the input winding  191   b  to set the number of turns of the cancellation winding  192   b  wound per unit area of one winding layer to set the amount of capacitive coupling to the output winding  193  to a target value. 
     In the transformer  19   b  of the present invention according to Principle  1  in  FIG. 7 , the winding layer  191   b - c  having the highest potential variation among the winding layers of the input winding  191   b  may be located between the winding layer  191   b - a  having the lowest potential variation and the remaining layer  191   b - b  of the input winding. 
     In the transformer  19   b  of the present invention according to Principle  1  in  FIG. 7 , the number of turns of a winding layer located at an end in the opposite direction to the direction facing the output winding  193  among the winding layers of the input winding  191   b  may be selected to be different from the number of turns of other winding layer of the input winding ( 191   a  or  191   b - a  to  191   b - c ) to set the number of turns of the cancellation winding  192   b  wound per unit area of one winding layer to a target value. 
     Principle  2  in  FIG. 12  proposes a method of setting the strength of an electric field formed in the opposite direction to the direction facing the output winding  193  regardless of the potential variation width of the  191   c.    
     The input winding  191   c  and output winding  193  in  FIG. 12  correspond to the input winding  191   a  and output winding  193  in  FIG. 7 . 
     In  FIG. 12 , the core bias winding  194  having a potential variation with the same polarity as the potential variation of the input winding  191   c  shields a capacitive coupling between a layer having a high potential variation of the input winding  191   c  and the core  196  of the transformer  19   c  as well as forms an electric field in the opposite direction to the direction facing the output winding  193  due to a potential contained in the input winding  191   c  and the core bias winding  194  to capacitively couple the transformer core  196  and capacitively couple the output winding  193  through a magnetic path of the core. 
     The cancellation winding  192   c  fills one winding layer between the input winding  191   c  and the output winding  193  with no gap to shield a capacitive coupling due to an electric field generated in the direction of the input winding  191   c  facing the output winding  193 , and cancels out and removes a minute coupling current generated in spite of shielding and a coupling current due to an electric field generated in the direction of the input winding  191   c  and the core bias winding  194  facing the output winding  193  with a capacitive coupling current generated between the cancellation winding  192   a  and the output winding  193 . 
     Accordingly, when the number of turns of the core bias winding  194  is selected in an appropriate manner, the number of turns of the cancellation winding  192   c  required for cancellation can be set to a value suitable to the productivity. 
       FIG. 13  is an embodiment illustrating a transformer configured according to Principle  2  in  FIG. 12 . 
     In the transformer  19   c  of  FIG. 13 , a winding layer having the lowest potential variation among the winding layers of the input winding  191   c  is wound closest to the output winding  193 , and a winding layer having the highest potential variation is located farthest from the output winding  193 . The other description thereof is the same as the description of  FIG. 12 . 
       FIG. 14  is a configuration diagram illustrating a flyback converter to which the transformer  19   c  is applied, and the elements other than the transformer  19   c  correspond to those of  FIG. 9 . 
     The present invention described above with reference to  FIGS. 12 through 14  will be summarized again below. 
     The transformer  19   c  of the present invention according to Principle  2  in  FIG. 12  may further include a core bias winding  194  wound between a winding layer located farthest from the output winding  193  among the winding layers of the input winding  191   c  and the core  196  of the transformer to have the same polarity of potential variation as that of potential variation at a connecting point between an end of the input winding  191   c  and an end of the switching element  12 , wherein an amount of capacitive coupling generated to the output winding  193  by an electric field formed in the opposition direction to the direction facing the output winding  193  from the winding surface of a winding layer located at an opposite end in the direction of the input winding  191   c  facing the output winding  193  is set by the number of turns of the core bias winding  194 . 
       FIG. 15  is a modification of Principle  2  in  FIG. 12 . 
     The transformer  19   c  of  FIG. 12  generates an electric field in the opposite direction to the direction facing the output winding  193  by the core bias winding  194 . On the contrary, the transformer  19   d  of  FIG. 15  allows a terminal end of the core bias winding  194   d  to be directly connected to the transformer core  196 , thereby allowing the transformer core  196  to generate an electric field by a potential of the core bias winding  194 . The input winding  191   d , output winding  193 , and cancellation winding  192   d  in  FIG. 15  correspond to the input winding  191   c , output winding  193 , and cancellation winding  192   c  in  FIG. 12 , respectively. 
     The cancellation winding  192   d  of  FIG. 15  shields a capacitive coupling due to an electric field generated in the direction of the input winding  191   d  facing the output winding  193 , and cancels out and removes a minute coupling current generated in spite of shielding and a coupling current due to an electric field generated from the transformer core  196  with a capacitive coupling current due to a potential difference between the cancellation winding  192   d  and the output winding  193 . 
     Accordingly, the number of turns of the cancellation winding  192   d  required for cancellation can be set to its desired value by appropriately selecting the number of turns of the core bias winding  194   d.    
       FIG. 16  is a configuration diagram illustrating a flyback converter to which the transformer  19   d  of  FIG. 15  is applied, and the elements other than the transformer  19   c  correspond to those of  FIG. 9 . 
     The present invention described above with reference to  FIGS. 15 and 16  will be summarized again below. 
     The transformer  19   d  of the present invention according to Principle  2  in  FIG. 12  may further include a core bias winding  194   d  wound between a winding layer located farthest from the output winding  193  among the winding layers of the input winding  191   d  and the transformer core  196 , in which one side terminal with the same polarity of potential variation as that of potential variation at a connecting point between an end of the input winding  191   d  and an end of the switching element  12  is connected to the transformer core  196 , wherein an amount of capacitive coupling generated to the output winding  193  by an electric field formed from the transformer core  196  is set by the number of turns of the core bias winding  194   d.    
     As described above, the present invention may set an amount of capacitive coupling to the output winding  193  through the transformer core  196  and the like by an electric field generated in the opposite direction to the direction facing the output winding  193 , thereby allowing the number of turns of the cancellation winding  192   a  to  192   d  for reducing a displacement current flowing to the electrical ground from the power supply to be set to the number of turns for good productivity and suitable to take out auxiliary power. 
     Furthermore, the cancellation winding  192   a  to  192   d  of the transformer  19   a  to  19   d  according to the present invention may facilitate the winding work to enhance the productivity, and the physical location variation of the cancellation winding  192   a  to  192   d  filled in one winding layer with no gap by a large number of turns may be small to generate a low deviation of capacitive coupling to the output winding, and as a result, a deviation of the cancellation characteristics may be generated to a small extent, thereby stabilizing the deviation of conducted EMI to a large extent even during mass production, and having an effect on cost reduction due to low unit production cost. 
     [Second Embodiment] 
     In the transformer  13   a  in the related art and the transformer  19   a  to  19   d  in  FIGS. 8 through 16 , the cancellation winding  132  or  192   a  to  192   d  deteriorates a magnetic coupling between the input winding  131  or  191   a  to  191   d  and the output winding  133  or  193  to increase a leakage inductance and deteriorate the efficiency. Furthermore, the cancellation winding  132  or  192   a  to  192   d  cancels out a capacitive coupling between the input winding  131  or  191   a  to  191   d  and the output winding  133  or  193  using an induced voltage, wherein the induced voltage may have a delayed distorted waveform compared to a voltage waveform of the input winding  131  or  191   a  to  191   d  and thus the cancellation effect may vary for each frequency bandwidth due to an error of cancellation. 
       FIG. 17  illustrates Principle  3  proposing a solution of increasing a magnetic coupling between the input winding and the output winding and reducing a leakage inductance to enhance the efficiency, and reducing an error of cancellation to provide an excellent cancellation effect. 
     Referring to  FIG. 17 , the transformer core  226 , first input winding  221   a , and output winding  223  correspond to the transformer core  196 , input winding  191   a , and output winding  193  in  FIG. 7 , and the cancellation winding  192   a  in  FIG. 7  is replaced with the second input winding  222   a.    
     Similarly to the description of  FIG. 7 , a capacitive coupling between the first input winding  221   a  and the output winding  223  is made of a coupling due to an electric field generated in the direction of the first input winding  221   a  facing the output winding  223  and a coupling due to an electric field generated in the opposite direction to the direction of the first input winding  221   a  facing the output winding  223  as illustrated in  FIG. 17 . The second input winding  222   a  is wound to fill one winding layer between the input winding  221   a  and the output winding  223  with no gap to shield a capacitive coupling due to an electric field generated in the direction of the input winding  221   a  facing the output winding  223 . 
     A minute coupling current generated through a surface facing each other between the first input winding  221   a  and the output winding  223  in spite of shielding and a coupling current through the transformer core  226  due to an electric field generated in the opposite direction to the direction of the input winding  221   a  facing the output winding  223  are cancelled out and reduced with a capacitive coupling current between the second input winding  222   a  and the output winding  223 . Furthermore, similarly to the description of the cancellation winding  192   a  in  FIG. 7 , the number of turns of the second input winding  222   a  required for cancellation is greater than that of the output winding  223 . 
       FIG. 18  illustrates a transformer  22   a  to which Principle  3  in  FIG. 17  is applied, and  FIG. 19  is a flyback converter to which the transformer  22   a  of  FIG. 18  is applied, and  FIGS. 18 and 19  will be described below. 
     The input winding wound around the transformer core  226  of the transformer  22   a  is divided into the first input winding  221   a  and the second input winding  222   a . As illustrated in  FIG. 19 , the first input winding  221   a  connected between a “+” input voltage and the switching element  12  and the second input winding  222   a  connected between a “−” input voltage and the switching element  12  transfer magnetic energy with potential variation in opposite polarities to each other due to switching of the switching element  12  by the control of the driving circuit  18  to, and the output voltage of the output winding  223  is rectified and smoothened by the output rectifier  14   a  and capacitor  15  to supply energy to the load. As a portion of the input winding, the second input winding  222   a  transfers energy, thereby having an advantage in that the level of coupling to the output winding  223  is high to have a low leakage inductance, and the energy transfer efficiency is higher than the transformer  13   a  in the related art and the transformer  19   a  to  19   d  in the related art in  FIGS. 8 through 16 . 
     A potential variation or high frequency noise generated from the first input winding  221   a  has an opposite polarity to that of the potential variation or high frequency noise generated from the second input winding  222   a , and a amount transferred from the two windings to other elements and lines within the power supply is cancelled out and thus only the difference of amount remains. Consequently, if the variations of an electric field or the sizes of high frequency noise from the two windings are the same, then the value of noise transferred to other elements or lines within the power supply is cancelled out to become very low. 
     The first input winding  221   a  and second input winding  222   a  have the same current change according to the switching operation of the switching element  12 , and the two windings generate symmetrical waveforms with opposite polarities at the same instant. Accordingly, the voltage of the second input winding  222   a  has a much more similar waveform to the voltage waveform of the first input winding  221   a  compared to the cancellation winding  192   a  using an induced voltage in  FIG. 7 , and thus more accurate cancellation operation is enabled, thereby having an excellent cancellation effect over a broad frequency bandwidth. 
     Referring to  FIG. 18 , similarly to the description of  FIG. 7 , in order to allow the number of turns of the second input winding  222   a  required for cancellation to be greater than that of the output winding  223 , the voltage of a terminal of the output winding  223  connected to the output rectifier  14   a  is configured to have an opposite polarity to the potential variation of the first input winding  221   a . Accordingly, as illustrated in  FIG. 19 , the output voltage rectified through the output rectifier  14   a  and smoothened by the capacitor  15  is a “−” voltage. 
     Referring to  FIG. 18 , the second input winding  222   a  cancels out and removes the sum of capacitively couplings to the output winding  223  from windings other than the second input winding  222   a  and the transformer core  226  with a capacitive coupling generated between the second input winding  222   a  and the output winding  223 , thereby reducing a displacement current flowing to the electrical ground through the output line  17  of the power supply to a very small extent. 
     As illustrated in  FIG. 7  for the cancellation winding  192   a , in order to cancel out a capacitive coupling generated from the first input winding  221   a  to the output winding  223  having a potential with the opposite polarity, the second input winding  222   a  should have a potential with the opposite polarity greater than that of the output winding  223  and have a number of turns greater than that of the output winding  223 . Furthermore, the number of turns of the second input winding  222   a  for cancellation may be set far greater than that of the output winding  223  according to an amount of capacitive coupling due to an electric field generated in the opposite direction to the direction of the first input winding  221   a  facing the output winding  223 . Furthermore, the flyback voltage of the second input winding  222   a  may be rectified and smoothened by a diode  20  and a capacitor  22  and used as an auxiliary power source for the driving circuit  18 . In this case, additional windings for supplying the auxiliary power source may be not required, thus simplifying the structure of windings to reduce the cost. 
     Referring to  FIG. 18 , high frequency noises generated from the first input winding  221   a  and second input winding  222   a  by a current change of the switching element  12  illustrated in  FIG. 19  have symmetrically opposite polarities to each other. 
     When the first input winding  221   a  and second input winding  222   a  are capacitively coupled to allow high frequency noise generated from the first input winding  221   a  to be overlapped with the second input winding  222   a , high frequency noise of the second input winding  222   a  is cancelled out and reduced. In this case, low high frequency noise is transferred to the output winding  223  wound and capacitively coupled to face the second input winding  222   a , thereby having an additional advantage in that high frequency noise radiation through the output line of the power supply is reduced. Furthermore, high frequency noise generated from the second input winding  222   a  is overlapped with the first input winding  221   a  and thus high frequency noise generated from the first input winding  221   a  is cancelled out and weakened. 
     In the actual use, resistors and capacitors may be also placed at appropriate positions, such as the first input winding  221   a , second input winding  222   a , output winding  223 , switching element  12 , output rectifier  14   a , or the like to further reduce high frequency noise radiation, but it is generally known and thus not described in all the drawings proposed to describe the present invention. 
     The present invention described above with reference to  FIGS. 17 through 19  will be summarized again below. 
     The transformer  22   a  of the present invention according to Principle  3  in  FIG. 17  may include a core  226  of the magnetic energy-transfer element; a first input winding  221   a  wound around the core  226  of the transformer, and connected between the “+” input voltage terminal and one side terminal of the switching element  12 , wherein the flow of current and the transfer of magnetic energy are switched by the switching operation of the switching element  12 ; and a second input winding wound  222   a  around the core  226  of the transformer, and connected between the “−” input voltage terminal and the other side terminal of the switching element  12 , wherein the flow of current and the transfer of magnetic energy are switched by the switching operation of the switching element  12 , wherein an effect exerted to the outside due to a potential variation and generated noise of the first input winding  221   a  by the switching operation of the switching element  12  and an effect exerted to the outside due to a potential variation and generated noise of the second input winding  222   a  by the switching operation of the switching element  12  are cancelled out due to their opposite polarities. 
     Furthermore, the transformer  22   a  of the present invention according to Principle  3  in  FIG. 17  may include a core  226  of the magnetic energy-transfer element; a first input winding  221   a  wound around the core  226  of the transformer, and connected between the “+” input voltage terminal and one side terminal of the switching element  12 , wherein the flow of current and the transfer of magnetic energy are switched by the switching operation of the switching element  12 ; a second input winding wound  222   a  around the core  226  of the transformer, and connected between the “−” input voltage terminal and the other side terminal of the switching element  12 , wherein the flow of current and the transfer of magnetic energy are switched by the switching operation of the switching element  12 ; and an output winding  223  magnetically coupled to the first input winding  221   a  and the second input winding  222   a  to take out energy, wherein an effect exerted to the outside due to a potential variation and generated noise of the first input winding  221   a  by the switching operation of the switching element  12  and an effect exerted to the outside due to a potential variation and generated noise of the second input winding  222   a  by the switching operation of the switching element  12  are cancelled out due to their opposite polarities. 
     Furthermore, in the transformer  22   a  of the present invention according to Principle  3  in  FIG. 17 , high frequency noise generated and emitted from the first input winding  221   a  by the switching operation of the switching element  12  and high frequency noise generated and emitted from the second input winding  222   a  by the switching operation of the switching element  12  have opposite polarities and thus are cancelled out each other. 
     Furthermore, in the transformer  22   a  of the present invention according to Principle  3  in  FIG. 17 , as disclosed in the description of  FIG. 18 , a capacitive coupling generated to lines and elements within the power supply due to a potential variation of the first input winding  221   a  by the switching operation of the switching element  12  and a capacitive coupling with the opposite polarity generated to lines and elements within the power supply due to a potential variation of the second input winding  222   a  by the switching operation of the switching element  12  have opposite polarities and thus are cancelled out. 
     Furthermore, in the transformer  22   a  of the present invention according to Principle  3  in  FIG. 17 , the second input winding  222   a  is located between the first input winding  221   a  and the output winding  223 . 
     Furthermore, in the transformer  22   a  of the present invention according to Principle  3  in  FIG. 17 , in order to reduce the conducted noise of the power supply including the transformer  22   a , the number of turns of the second input winding  222   a  wound per unit area of one winding layer for generating a capacitive coupling between the second input winding  222   a  and the output winding  223  required to cancel out and reduce the sum of capacitive couplings generated from windings other than the output winding  223  and the core  226  of the transformer to the output winding  223  is greater than that of the output winding  223  wound per unit area of one winding layer. 
     Furthermore, in the transformer  22   a  of the present invention according to Principle  3  in  FIG. 17 , an electric field generated from a winding surface of the winding located at an opposite end in the direction of the first input winding  221   a  facing the output winding  223  to the opposite direction to the direction of the first input winding  221   a  facing the output winding  223  is capacitively coupled to the second input winding  222   a  through the core  226  of the transformer. 
     Furthermore, in the transformer  22   a  of the present invention according to Principle  3  in  FIG. 17 , as increasing an amount of capacitive coupling generated to the output winding  223  by an electric field formed in the opposition direction to the direction facing the output winding  223  from the winding surface of a winding layer located at an opposite end in the direction facing the output winding  223  among the winding layers of the first input winding  221   a , the number of turns of the second input winding  222   a  wound per unit area of one winding layer is greater than that of the output winding  223  wound per unit area of one winding layer. 
     Referring to  FIG. 18 , in the transformer  22   a , a winding layer having the highest potential variation among the winding layers of the input winding  221   a  is located at an end in the opposite direction to the direction facing the output winding  223  and generates an electric field in the opposite direction to the direction facing the output winding  223 . As illustrated in  FIG. 8 , when the input voltage is very high in the structure of the transformer  22   a , an electric field generated by a winding layer having the highest potential variation among the winding layers of the first input winding  221   a  is very high, and thus required to be reduced. 
       FIG. 20  illustrates the structure of a transformer  22   b  which is an embodiment corresponding to a case of high input voltage. In the transformer  22   b  of  FIG. 20 , a winding layer  221   b - a  having the lowest potential variation among the winding layers of the first input winding  221   b  is located at an end in the opposite direction to the direction facing the output winding  223  and generates an electric field in the opposite direction to the direction facing the output winding  223 . Furthermore, in the transformer  22   b , a winding layer  221   b - c  having the highest potential variation among the winding layers of the first input winding  221   b  is located between the winding layer  221   b - a  having the lowest potential variation and a winding layer  221   b - b  having a middle potential variation, and thus it is prevented for a high spike voltage of the winding layer  221   b - c  having the highest potential variation to be coupled capacitively to the second input winding  222   b , thus distorting the waveform and a cancellation error does not occur. 
     As disclosed in the description of  FIG. 10 , the location arrangement of a winding layer having the lowest potential variation width, a winding layer having the highest potential variation width, and a winding layer having a middle potential variation width among the winding layers of the first input winding  221   b - a  to  221   b - c  may be configured in various ways according to the size of the input voltage or the size of the potential variation width of the first input winding  221   b.    
     The transformer  22   b  generates an electric field in the opposite direction to the direction facing the output winding  223  by the potential of a winding layer located at an end in the opposite direction to the direction facing the output winding  223 . The strength of an electric field generated in the opposite direction to the direction facing the output winding  223  can be set by choosing the location and number of turns of each winding layer  221   b - a  to  221   b - c  of the first input winding  221   b , and the number of turns of the second input winding  222   b  for cancellation may be set to a desired value suitable to the productivity. 
     On the other hand, in the transformer  22   b , the high frequency noise of the winding layer  221   b - b  having a middle potential variation among the winding layers of the first input winding  221   b - a  to  221   b - c  is transferred to the second input winding  222   b  through a distributed capacitance to be overlapped, thereby cancelling out and reducing high frequency noise generated from the second input winding  222   b.    
       FIG. 21  is a configuration diagram illustrating a flyback transformer to which the transformer  22   b  is applied, and the elements other than the transformer  22   b  correspond to those of  FIG. 19 . 
     The present invention described above with reference to  FIGS. 20 and 21  will be summarized again below. 
     In the transformer  22   b  of the present invention according to Principle  3  in  FIG. 17 , the location arrangement of the winding layer  221   b - a  having the lowest potential variation width, the winding layer  221   b - b  having the highest potential variation width, and winding layer  221   b - c  having a middle potential variation width among the winding layers of the first input winding  221   b  may be configured in various ways. 
     Furthermore, in the transformer  22   b  of the present invention according to Principle  3  in  FIG. 17 , the winding layer  221   b - b  having the highest potential variation width among the winding layers of the input winding  221   b  may be located between the winding layer  221   b - a  having the lowest potential variation width among the winding layers of the input winding  221   b  and the remaining winding layers. 
     Furthermore, in the transformer  22   b  of the present invention according to Principle  3  in  FIG. 17 , the number of turns of a winding layer located at an end in the opposite direction to the direction facing the output winding  223  among the winding layers of the first input winding  221   b - a  to  221   b - c  may be differently selected from the number of turns of the other winding layers of the first input winding  221   b - a  to  221   b - c  to set the number of turns of the second input winding  222   b  wound per unit area of one winding layer to a target value. 
       FIG. 22  illustrates the structure of a transformer  22   c  in which the strength of an electric field generated in the opposite direction to the direction facing the output winding  223  is set by the core bias winding  224 , thus setting the number of turns of the second input winding  222   c  for cancelling out a capacitive coupling generated to the output winding  223  to a desired value, as disclosed in the description of  FIGS. 12 and 13 . 
     Referring to  FIG. 22 , when the number of turns of the second input winding  222   c  required for cancellation is not required to be greater than that of the output winding  223 , the number of turns of the core bias winding  224  may be small, and under the circumstances, the potential variation of the core bias winding  224  has an opposite polarity to that of the potential variation of the first input winding  221   c , and thus the core bias winding  224  may be used for the purpose of shielding a capacitive coupling between the first input winding  221   c  and the transformer core  226 . 
       FIG. 23  is a configuration diagram illustrating a flyback converter to which the transformer  22   c  is applied, and illustrates an embodiment in which the sum of flyback voltages of the core bias winding  224  and second input winding  222   c  of the transformer  22   c  is rectified and smoothened with the capacitor  31  and used as an auxiliary power source for the driving circuit  18 . The other elements correspond to those of  FIG. 19 . 
     The present invention described above with reference to  FIGS. 22 and 23  will be summarized again below. 
     The transformer  22   c  of the present invention according to Principle  3  in  FIG. 17  may further include a core bias winding  224  configured to shield a capacitive coupling due to a potential variation between a winding layer located farthest from the output winding  223  among the winding layers of the first input winding  221   c  and the transformer core  226 . The core bias winding  224  may have a potential variation with the same polarity or opposite polarity as that of the potential variation of the first input winding  221   c.    
     Furthermore, the transformer  22   c  of the present invention according to Principle  3  in  FIG. 17  may further include a core bias winding  224  wound between a winding layer located farthest from the output winding  223  among the winding layers of the first input winding  221   c  and the transformer core  226  to have the same polarity of potential variation as that of potential variation of the first input winding  221   c , wherein an amount of capacitive coupling generated to the output winding  223  by an electric field formed from a winding surface of the winding layer located at an opposite end in the direction of the first input winding  221   c  facing the output winding  223  in the opposite direction to the direction facing the output winding  223  is set by the number of turns of the core bias winding  224 . 
       FIG. 24  illustrates a transformer  22   d  in which a second input winding  222   d  is located between a first input winding  221   d  and the output winding  223  and a cancellation winding  225  is located between the second input winding  222   d  and the output winding  223 . 
     The transformer  22   d  may be used to prevent an affect caused by the application of a surge voltage such as static electricity applied to the output winding  223 , and the surge voltage transferred to the cancellation winding  225  through the output winding  223  is bypassed to the AC ground and thus a reduced voltage is applied to the second input winding  222   d  to protect the switching element  12  or the like. 
     The cancellation winding  225  of the transformer  22   d  is wound to fill one winding layer between the second input winding  222   d  and the output winding  223  with no gap and shield a capacitive coupling due to an electric field generated in the direction of the first input winding  221   d  and second input winding  222   d  facing the output winding  223 , thus generating a very low electric field. 
     The cancellation winding  225  of the transformer  22   d  generates a capacitive coupling between the cancellation winding  225  and the output winding  223  to cancel out and remove the sum of capacitive couplings generated from windings other than the cancellation winding  225  and the transformer core  226  contained in the transformer  22   d  to the output winding  223 . The number of turns of the cancellation winding  225  for cancellation should have a potential variation with the greater opposite polarity than that of the output winding  223  to generate a capacitive coupling with the opposite polarity to the sum of capacitive couplings among the output winding  223  having a potential variation with the opposite polarity to that of the potential variation of the input winding  221   d  and the first input winding  221   d  and second input winding  222   d . To this end, the number of turns of the cancellation winding  225  is greater than that of the output winding  223 . 
       FIG. 25  is a configuration diagram illustrating a flyback converter to which the transformer  22   d  is applied, and the elements other than the transformer  22   d  correspond to those of  FIG. 19 . 
     The present invention described above with reference to  FIGS. 24 and 25  will be summarized again below. 
     The transformer  22   d  of the present invention according to Principle  3  in  FIG. 17  may further include a cancellation winding  225  configured to shield a capacitive coupling due to a potential variation between the second input winding  222   d  and the output winding  223 , and cancel out the sum of capacitive couplings due to a potential variation generated from windings other than the output winding  223  and the transformer core  226  to the output winding. 
     Furthermore, in the transformer  22   d  of the present invention according to Principle  3  in  FIG. 17 , the number of turns of the cancellation winding  225  wound per unit area of one winding layer required to cancel out and reduce the sum of capacitive couplings due to a potential variation generated from windings other than the output winding  223  and the transformer core  226  to the output winding is greater than that of the output winding  223  wound per unit area of one winding layer. 
       FIG. 26  illustrates a transformer  22   e  in which a terminal end of the core bias winding  224   e  is connected to the transformer core  226  to allow the transformer core  226  to form an electric field with a potential of the core bias winding  224   e , thereby determining an amount of capacitive coupling to the output winding  223 , and setting the number of turns of the second input winding  222   e  required for cancellation to a desired value by controlling the number of turns of the core bias winding  224   e , and  FIG. 27  is a configuration diagram illustrating a flyback converter to which the transformer  22   e  is applied, and the elements other than the transformer  22   e  correspond to those of  FIG. 19 . 
     The present invention described above with reference to  FIGS. 24 and 25  will be summarized again below. 
     The transformer  22   e  of the present invention according to Principle  3  in  FIG. 17  may further include a core bias winding  224   e  wound between a winding layer located farthest from the output winding  223  among the winding layers of the first input winding  221   e  and the transformer core  226  and one side terminal having a potential variation with the same polarity as a potential variation of the first input winding  221   e  is connected to the transformer core  226 , wherein an amount of capacitive coupling generated to the output winding  223  by an electric field formed in the transformer core  226  is set by the number of turns of the core bias winding  224   e.    
       FIGS. 19, 21, 28 through 30  are configuration diagrams illustrating a flyback converter of the present invention for suppressing the generation of high frequency noise or preventing high frequency noise from being transferred to the output winding. 
     Referring to  FIG. 19 , high frequency noise generated from the first input winding  221   a  due to a fast change of current flow by driving the switching element  12  has an opposite polarity to high frequency noise generated from the second input winding  222   a  having the same change of current flow. The first input winding  221   a  and the second input winding  222   a  are capacitively coupled through a distributed capacitance, and high frequency noise with the opposite polarity generated from the first input winding  221   a  is overlapped with high frequency noise generated from the second input winding  222   a , and thus the high frequency noise of the second input winding  222   a  is cancelled out and reduced. The reduced high frequency noise is transferred to the output winding  223  located to face the second input winding  222   a  having the reduced high frequency noise, and thus high frequency noise radiation through the output line of the power supply is reduced. Furthermore, high frequency noise with the opposite polarity generated from the second input winding  222   a  is also transferred to the first input winding  221   a , and high frequency noise generated from the second input winding  222   a  is overlapped with high frequency noise generated from the first input winding  221   a , and thus the high frequency noise of the first input winding  221   a  is cancelled out and reduced. 
     The present invention described above with reference to  FIG. 19  will be summarized again below. 
     In the transformer  22   a  of the present invention according to Principle  3  in  FIG. 17 , the first input winding  221   a  and the second input winding  222   a  are capacitively coupled, and thus high frequency noise generated from the first input winding  221   a  and high frequency noise with the opposite polarity generated from the second input winding  222   a  are cancelled out and reduced. 
     Furthermore, in the transformer  22   a  of the present invention according to Principle  3  in  FIG. 17 , the first input winding  221   a  and the second input winding  222   a  are capacitively coupled, and thus high frequency noise generated from the second input winding  222   a  is cancelled out and reduced by high frequency noise with the opposite polarity generated from the first input winding  221   a , and high frequency noise transferred from the second input winding  222   a  to the output winding  223  is reduced. 
     Referring to  FIG. 19 , in order to cancel out and remove high frequency noise generated from the second input winding  222   a , the size of noise transferred from the first input winding  221   a  to the second input winding  222   a  should be the same as the size of high frequency noise generated from the second input winding  222   a . To this end, a winding layer of the first input winding  221   a  capacitively coupled to the second input winding  222   a  among the winding layers of the first input winding  221   a  through a distributed capacitance to transfer noise with the same size as that of high frequency noise generated from the second input winding  222   a  should be selected and coupled thereto. 
       FIG. 21  is one of solutions for that purpose, illustrating an example in which a middle layer  221   b - b  of the first input winding  221   b  and the second input winding  222   b  are coupled by a distributed capacitance between windings, and thus high frequency noise generated from the second input winding  222   b  is cancelled out and removed by high frequency noise with the opposite polarity generated from the middle layer  221   b - b  of the first input winding  221   b . A ratio of the number of turns of each layer of the first input winding  221   b  is chosen to set the size of high frequency noise of the middle layer  221   b - b  of the first input winding  221   b  to a value required to remove the high frequency noise of the second input winding  222   b.    
     Referring to  FIG. 21 , as decreasing the ratio of the number of turns of the second input winding  222   b  with respect to the number of turns of the first input winding  221   b , the size of high frequency noise generated from the second input winding  222   b  is less than that generated from the first input winding  221   b , and greater than that generated from the first input winding  221   b  as increasing the ratio. 
     Accordingly, one of the winding layers  221   b - a  to  221   b - c  of the first input winding  221   b  coupled to the second input winding  222   b  should be selected to cancel out and remove high frequency noise generated from the second input winding  222   b  according to the ratio of the number of turns of the second input winding  222   b  with respect to the number of turns of the first input winding  221   b . Furthermore, in order to remove the high frequency noise of the second input winding  222   b , the number of turns of each winding layer  221   b - a  to  221   b - c  of the first input winding  221   b  may be chosen, and the size of the overlapped high frequency noise with the second input winding  222   b  through a distributed capacitance between windings and the size of the generated high frequency noise with the opposite polarity from the second input winding  222   b  are able to set to be equal. 
     The present invention described above with reference to  FIG. 21  will be summarized again below. 
     In the transformer  22   b  of the present invention according to Principle  3  in  FIG. 17 , the first input winding  221   b  and the second input winding  222   b  are coupled through a distributed capacitance between the two windings and thus high frequency noise generated from the second input winding  222   b  is cancelled and reduced by high frequency noise with the opposite polarity generated from the first input winding  221   b.    
     In the transformer  22   b  of the present invention according to Principle  3  in  FIG. 17 , a winding layer of the first input winding  221   b  located closest to the second input winding  222   b  is one of the winding layer  221   b - a  having the lowest potential variation width, the winding layer  221   b - c  having the highest potential variation width, and the winding layer  221   b - b  having a middle potential variation width. 
     In the transformer  22   b  of the present invention according to Principle  3  in  FIG. 17 , the number of turns of one or more winding layers of the first input winding  221   b  may be differently selected from that of the other winding layers to configure the size of high frequency noise of the first input winding  221   b  coupled to the second input winding  222   b  through a distributed capacitance to an optimal size required for cancellation. 
       FIG. 28  illustrate an example in which part of the first input winding  221   b  and the second input winding  222   b  are coupled through the resistor  24  and capacitor  23  in addition to coupling the middle layer  221   b - b  of the first input winding  221   b  and second input winding  222   b  through a distributed capacitance between the windings. Noise transferred from the first input winding  221   b  due to coupling cancels and removes high frequency noise with the opposite polarity generated from the second input winding  222   b , thereby preventing high frequency noise from being transferred to the output winding  223 . 
       FIG. 29  is an embodiment in which a tap  251   a  and  251   b  of the first input winding  251  and the second input winding  252  are coupled through the resistor  24  and capacitor  23 , in order to cancel out the high frequency noise of a partial winding of the first input winding  251  and the high frequency noise with the opposite polarity of the second input winding  252  in parallel with the cancellation of high frequency noise through a distributed capacitance between a winding layer of the first input winding  251  and the second input winding  252  in a typical winding structure. 
     The present invention described above with reference to  FIGS. 28 and 29  will be summarized again below. 
     In the transformer  22   b  or  25  of the present invention according to Principle  3  in  FIG. 17 , as disclosed in the description of  FIGS. 19, 21, 28 and 29 , the high frequency noise generated from the second input winding  222   b  or  225  is cancelled out and reduced by high frequency noise with the opposite polarity generated from the first input winding  221   b  or  251 - a  and  251 - b  by a coupling through a distributed capacitance between the two windings of the first input winding  221   b  or  251 - a  and  251 - b  and the second input winding  222   b  or  252  and a capacitive coupling through one or more coupling elements. The coupling element may be the capacitor  23  or the capacitor  23  and resistor  24 . 
     The capacitor of  FIG. 28  requires high voltage capability, and is a high cost component.  FIG. 30  is provided to change or remove it to a low cost component. 
     Referring to  FIG. 30 , the first coupling winding  264  is coupled to part of the first input winding  261  through a distributed capacitance, and the first coupling winding  264  is connected to the second input winding  262  through the capacitor  23  and resistor  24 , and high frequency noise generated from part of the first input winding  261  is transferred to the first coupling winding  264  through a distributed capacitance and overlapped with the high frequency noise with the opposite polarity of the second input winding  262  to cancel out the high frequency noise of the second input winding  262 . Here, the first coupling winding  264  may be connected to the second input winding  262  through the resistor  24  or directly connected to the second input winding  262 . Referring to  FIG. 30 , when the high frequency noise of the second input winding  262  is effectively removed, the first coupling winding  264  does not have high frequency noise, and the first input winding  261  does not radiate high frequency noise to the outside because of surrounding with the winding  264  and second input winding  262  from which high frequency noise is removed. 
     The present invention described above with reference to  FIG. 30  will be summarized again below. 
     The transformer  26  of the present invention according to Principle  3  in  FIG. 17  may further include the first coupling winding  264  wound to face the first input winding  261  as disclosed in the description of  FIG. 30 , and the first input winding  261  and second input winding  262  are coupled through a distributed capacitance between the first input winding  261  and the second input winding  262 , and also coupled through a distributed capacitance between the first input winding  261  and first coupling winding  264 , and thus high frequency noise generated from second input winding  262  is cancelled out and reduced by high frequency noise with the opposite polarity generated from the first input winding  261 . Here, the second input winding  262  and first coupling winding  264  may be connected to each other directly, or connected through the capacitor  23 , or through the capacitor  23  and resistor  24 , or through the resistor  24 . 
     According to the foregoing embodiments of the present invention, the number of turns of the second input winding  222   a  to  222   e  can be set to a desired value suitable to the productivity far greater than that of the output winding  223 , and one winding layer can be filled and wound with no gap with about two strands of thin wire, thereby enhancing the productivity in the winding work of the transformer. Furthermore, a flyback voltage of the second input winding  222   a  to  222   e  is rectified to supply an auxiliary power source, and thus the auxiliary winding may be removed compared to the related art in which the auxiliary winding should be separately wound, thereby reducing the unit cost of the transformer. Furthermore, a variation in the physical location of the second input winding  222   a  to  222   e  that fills one layer with no gap is low to generate a small deviation of coupling to the output winding, thereby stabilizing the deviation of EMI to a large extent due to uniform cancellation even during mass production. Furthermore, the generation and radiation of high frequency noise can be reduced, thereby reducing the cost of the line filter or the like. 
     [Third Embodiment] 
       FIG. 31  illustrates a transformer  27   a  that is an embodiment of a sandwich winding structure having a structure for cancelling out conducted noise and high frequency radiated noise according to the present invention, and  FIG. 32  is a configuration diagram illustrating a flyback converter to which the transformer  27   a  of  FIG. 31  is applied. 
     Referring to  FIGS. 31 and 32 , the input winding of the transformer  27   a  is divided into a first input winding  271  and a second input winding  272 . The first input winding  271  connected between a “+” input voltage and the switching element  12  and the second input winding  272  connected between a “−” input voltage and the switching element  12  store and emit magnetic energy with potential variations having the opposite polarity, respectively, by the switching operation of the switching element  12 , and thus deliver rectified and smoothened energy with the output rectifier  14   a  and capacitor  15  through the output winding  273  to the load. 
     The potential variation of a terminal connected to the switching element  12  among the terminals of the first input winding  271  of the transformer  27   a  is generated in an opposite polarity to that of a terminal connected to the switching element  12  among the terminals of the second input winding  272 , and high frequency noise generated from the first input winding  271  due to the same change of current flow by driving the switching element  12  has an opposite polarity to high frequency noise generated from the second input winding  272 . Accordingly, a coupling generated by a capacitive coupling to the input line  16  or output winding  273  due to a potential variation of the first input winding  271  has an opposite polarity to a coupling generated by a capacitive coupling to the input line  16  or output winding  273  due to a potential variation of the second input winding  272  and thus cancelled out, and a current flowing to the electrical ground through the input line  16  or output line  17  of the power supply is reduced to a large extent compared to  FIG. 1 . Furthermore, high frequency noise generated from the first input winding  271  and transferred to the input line  16  or output winding  273  is cancelled out by high frequency noise with the opposite polarity generated from the second input winding  272  and transferred to the input line  16  or output winding  273 , and thus radiated noise through the input line  16  or output line  17  is also reduced to a large extent compared to the related art in  FIGS. 1 through 6 . 
     Furthermore, a sufficiently large number of turns of the second input winding  272  can be taken compared to that of the output winding  273 , and thus it is easy to fill and wind one layer with one or two strands of thin wire, and moreover, a flyback voltage of the second input winding  272  can be rectified and smoothened with the diode  30  and capacitor  31  to take out an auxiliary power source voltage, and thus an auxiliary winding for taking out an additional auxiliary voltage may be not required. 
     In the transformer  27   a  of  FIG. 31 , both a winding layer  271   b  of the first input winding  271  and the second input winding  272  have a far lower potential variation compared to a high potential variation of the second input winding  131   b  in  FIG. 6 , and even if it has a sandwich winding structure, a large capacitive coupling as disclosed in the related art of  FIG. 6  does not occur. 
     In the transformer  27   a  which is an embodiment having a sandwich winding structure, a capacitive coupling between the output winding  273  and the first input winding  271  and a capacitive coupling between the output winding  273  and the second input winding  272  are cancelled out each other and removed. 
     For example, when the number of turns of the second input winding  272  is 30T and the number of turns of the output winding  273  is 8T in the same direction, it has a potential difference due to a difference of 22T in the number of turns with the same polarity between the output winding  273  and the second input winding  272 , and when the number of turns of the first input winding  271   b  capacitively coupled to face the output winding  273  among the winding layers of the first input winding  271  becomes 14T in the opposite direction, it also has a potential difference due to a difference of 22T in the number of turns. An electric field due to a potential of the winding layer  271   a  of the first input winding  271  is shielded by the winding layer  271   b  of the first input winding  271  wound to fill one winding layer with no gap, but by taking a capacitive coupling generated from the winding layer  271   a  of the first input winding  271  to the output winding  273  in spite of shielding into consideration, by increasing the number of turns of the second input winding  272  by 1T or 2T or reducing the number of turns of the winding layer  271   b  of the first input winding  271  by 1T or 2T, the sum of capacitive couplings generated from the first input winding  271  and second input winding  272  to the output winding  273  is are cancelled out and removed. 
     Furthermore, high frequency noise generated from the second input winding  272  of 30T and transferred to the input line  16  or output winding  273  is cancelled out by high frequency noise generated from the winding layer  271   a  having a high potential variation and the winding layer  271   b  of 14T and transferred to the input line  16  or output winding  273 , and thus radiated noise through the input line  16  or output line  17  is greatly reduced compared to the related art in  FIGS. 1 through 6 . 
     The output rectifier  14   a  of  FIG. 32  rectifies a negative voltage and smoothens it with the capacitor  15  to obtain a negative output voltage from the output winding  273  since a potential variation of the second input winding  272  of the transformer  27   a  and a potential variation of the output winding  273  have the same polarity. If the potential variation of the output winding  273  has the same polarity as that of the first input winding  271 , the direction of the output rectifier  14   a  is changed, and the output voltage obtained by smoothening with the capacitor  15  becomes a positive voltage. 
     The present invention described above with reference to  FIGS. 31 and 32  will be summarized again below. 
     In the transformer  27   a  of  FIG. 31 , the output winding  273  is located between the first input winding  271  and the second input winding  272 . 
     In the transformer  27   a  of  FIG. 31 , a capacitive coupling generated from the first input winding  271  to the output winding  273  and a capacitive coupling generated from the second input winding  272  to the output winding  273  are cancelled out and reduced. 
     The transformer  27   b  of  FIG. 33  may include a first shielding winding  274  between the first input winding  271  and the output winding  273  and a second shielding winding  275  between the second input winding  272  and the output winding  273  in a sandwich structure of the first input winding  271 , output winding  273  and second input winding  272 . 
     According to an example of the transformer  27   a  of  FIG. 31 , there is disclosed an example in which the second input winding  272  capacitively coupled to the output winding  273  has 30T and the winding layer  271   b  of the first input winding  271  capacitively coupled to the output winding  273  has 14T. High frequency noise generated from the second input winding  272  of the transformer  27   a  and transferred to the output winding  273  is different in size from that of high frequency noise generated from the winding layer  271   b  of the first input winding  271  and transferred to the output winding  273  though it is lower than the related art, and thus they are not completely cancelled out. Furthermore, there is a potential difference corresponding to 22T between the output winding  273  of 8T and the second input winding  272  of 30T, and thus an amount of the generated capacitive coupling is large, and even if removed through cancellation, there is a limit in reducing conducted noise through the output line  17 . 
       FIG. 33  provides a solution which effectively cancels out and removes the high frequency noise transferred from the first input winding  271  to the output winding  273  and the high frequency noise transferred from the second input winding  272  to the output winding  273  by setting both noise size equal, and as well as drastically reducing an amount of the generated capacitive coupling. 
     Referring to  FIG. 33 , the first shielding winding  274  and second shielding winding  275  of the transformer  27   b  shields a capacitive coupling generated from the first input winding  271  and second input winding  272  to the output winding  273 , and a coupling generated in spite of shielding is cancelled out by a capacitive coupling between the first shielding winding  274  and the output winding  273  and a capacitive coupling between the second shielding winding  275  and the output winding  273 . Furthermore, the number of turns of a layer wound closest to the output winding  273  among the winding layers of the first input winding  271  and second input winding  272  may be chosen to set the size of high frequency noise generated from the second input winding  272  and transferred to the output winding  273  to be same as that of high frequency noise with the opposite polarity generated from the first input winding  271  and transferred to the output winding  273 , and thus most of high frequency noise transferred to the output winding  273  is cancelled out and removed, thereby further reducing radiated noise through the output line  17  compared to the example of  FIG. 31 . 
     The present invention described above with reference to  FIG. 33  will be summarized again below. 
     The transformer  27   b  of  FIG. 33  may further include the first shielding winding  274  for shielding a capacitive coupling due to a potential variation between the first input winding  271  and the output winding  273 , and the second shielding winding  275  for shielding a capacitive coupling due to a potential variation between the second input winding  272  and the output winding  273 , in addition to the transformer  27   a  of  FIG. 31 . 
     The  27   c  of  FIG. 34  may include a second shielding winding  275  between the second input winding  272  and the output winding  273  in addition to the sandwich structure of the first input winding  271 , output winding  273  and second input winding  272  in  FIG. 31 . 
     In this case, the number of turns of the winding layer  271   b  located closest to the output winding  273  among the winding layers of the first input winding  271  may be chosen to be the same or similar to that of the second input winding  272 , and thus the size of high frequency noise generated from the second input winding  272  and transferred to the output winding  273  may be allowed to be identical to that of high frequency noise with the opposite polarity generated from the first input winding  271  and transferred to the output winding  273 . 
     The second shielding winding  275  shields a capacitive coupling generated from the second input winding  272  to the output winding  273 , and a coupling generated in spite of shielding and a capacitive coupling between the first input winding  271  and the output winding  273  are cancelled out by a capacitive coupling between the second shielding winding  275  and the output winding  273 . For example, when the number of turns of the winding layer  271   b  located closest to the output winding  273  among the winding layers of the first input winding  271  is 30T and the number of turns of the output winding  273  is 8T in the same direction, the number of turns of the second input winding  272  is chosen to about 30T to correspond to the size of high frequency noise. The second shielding winding  275  is chosen to about 14T in the opposite polarity to the first input winding  271  to generate a capacitive coupling with the same size but opposite polarity to a capacitive coupling due to a potential difference of 24T between the first input winding  271  and the output winding  273  to the output winding  273  for cancellation. 
     Accordingly, radiated noise through the output line  17  may be further reduced compared to the example of  FIG. 31 . 
     The present invention described above with reference to  FIG. 34  will be summarized again below. 
     The transformer  27   c  of  FIG. 34  may further include the second shielding winding  275  for shielding a capacitive coupling due to a potential variation between the second input winding  272  and the output winding  273 , in addition to the transformer  27   a  of the  FIG. 31 . 
     The  27   d  of  FIG. 35  may include a first shielding winding  274  between the first input winding  271  and the output winding  273  in (addition to) a sandwich structure of the first input winding  271 , output winding  273  and second input winding  272  of the  FIG. 31 . 
     Referring to  FIG. 35 , the number of turns of the winding layer  271   b  located closest to the output winding  273  among the winding layers of the first input winding  271  may be chosen to be the same or similar to that of the second input winding  272 , and thus the size of high frequency noise generated from the second input winding  272  and transferred to the output winding  273  may be allowed to be identical to that of high frequency noise with the opposite polarity generated from the first input winding  271  and transferred to the output winding  273 . 
     The first shielding winding  274  shields a capacitive coupling generated from the first input winding  271  to the output winding  273 , and a coupling generated in spite of shielding and a capacitive coupling between the second input winding  272  and the output winding  273  are cancelled out by a capacitive coupling between the first shielding winding  274  and the output winding  273 . For example, when the number of turns of the second input winding  272  is 30T and the number of turns of the output winding  273  is 8T in the same direction, the number of turns of the winding layer  271   b  located closest to the output winding  273  among the winding layers of the first input winding  271  is chosen to about 30T to correspond to the size of high frequency noise. The first shielding winding  274  is chosen to about 14T in the opposite polarity to the second input winding  272  to generate a capacitive coupling with the same size but opposite polarity to a capacitive coupling due to a potential difference of 24T between the second input winding  272  and the output winding  273  to the output winding  273  for cancellation. 
     The present invention described above with reference to  FIG. 35  will be summarized again below. 
     The transformer  27   d  of  FIG. 35  may further include the first shielding winding  274  for shielding a capacitive coupling due to a potential variation between the first input winding  271  and the output winding  273 , in addition to the transformer  27   a  of the  FIG. 31 . 
     Though not shown in the drawing, in the transformer  27   a  to  27   d  in  FIGS. 31 through 35 , the first input winding  271  and the second input winding  272  may be capacitively coupled using an additional coupling winding or external coupling element in  FIGS. 28 through 30  to cancel out noise generated from the first input winding  271  and second input winding  272 . 
     As an application example of  FIG. 30 , the transformer  27   a  to  27   d  may further include a first coupling winding wound to face part of the first input winding  271 , wherein part of the first input winding  271  and the second input winding  272  are coupled through a distributed capacitance between the first input winding  271  and the first coupling winding, thereby allowing high frequency noise generated from part of the first input winding  271  and high frequency noise with the opposite polarity generated from the second input winding  272  to be overlapped and cancelled out. 
     As another application example of  FIG. 30 , the transformer  27   a  to  27   d  may further include a second coupling winding wound to face the second input winding  272 , wherein the second input winding  272  and the first input winding  271  are coupled through a distributed capacitance between the second input winding  272  and the second coupling winding, thereby allowing high frequency noise generated from the first input winding  271  and high frequency noise with the opposite polarity generated from the second input winding  272  to be overlapped and cancelled out. 
     As still another application example of  FIG. 30 , the transformer  27   a  to  27   d  may further include a first coupling winding wound to face part of the first input winding  271  and a second coupling winding wound to face the second input winding  272 , wherein the second input winding  272  and the first input winding  271  are coupled through a distributed capacitance between the first input winding  271  and the first coupling winding and a distributed capacitance between the second input winding  272  and the second coupling winding, thereby allowing high frequency noise generated from the first input winding  271  and high frequency noise with the opposite polarity generated from the second input winding  272  to be overlapped and cancelled out. 
     The present invention not illustrated in the drawing will be summarized again below. 
     In the transformer  27   a  to  27   d  in  FIGS. 31 through 35 , as disclosed in the description of  FIGS. 28 through 30 , the first input winding  271  and the second input winding  272  are capacitively coupled to allow high frequency noise generated from the first input winding  271  to cancel out high frequency noise with the opposite polarity generated from the second input winding  272  as well as allow high frequency noise generated from the second input winding  272  to cancel out high frequency noise with the opposite polarity generated from the first input winding  271 , thereby reducing high frequency noise generated from the two windings. 
     Furthermore, in the transformer  27   a  to  27   d  in  FIGS. 31 through 35 , as disclosed in the description of  FIGS. 28 through 30 , the first input winding  271  and the second input winding  272  are capacitively coupled through one or more coupling elements, and thus high frequency noise generated from the first input winding  271  and high frequency noise with the opposite polarity generated from the second input winding  272  are cancelled out each other and reduced. The coupling element may be a capacitor or a capacitor and a resistor. A connecting point at which one side terminal of the coupling element is connected to the first input winding  271  is a connecting point between the first input winding  271  and the switching element  12  or a central tap of the first input winding  271 , and a connecting point at which the other side terminal of the coupling element is connected to the second input winding  272  is a connecting point between the second input winding  272  and the switching element  12  or a central tap of the second input winding  272 . 
     Furthermore, though not shown in the drawing, as an application example of  FIG. 30 , the transformer  27   a  to  27   d  in  FIGS. 31 through 35 , may further include a first coupling winding wound to face the first input winding  271 , and thus the first input winding  271  and the second input winding  272  are coupled through a distributed capacitance between the first input winding  271  and the first coupling winding, thereby cancelling out high frequency noise generated from part of the first input winding  271  and high frequency noise with the opposite polarity generated from the second input winding  272 . 
     Furthermore, though not shown in the drawing, as an application example of  FIG. 30 , the transformer  27   a  to  27   d  in  FIGS. 31 through 35  may further include a second coupling winding wound to face the second input winding  272 , and thus the second input winding  272  and the first input winding  271  are coupled through a distributed capacitance between the second input winding  272  and the second coupling winding, thereby cancelling out high frequency noise generated from the first input winding  271  and high frequency noise with the opposite polarity generated from the second input winding  272 . 
     Furthermore, though not shown in the drawing, as an application example of  FIG. 30 , the transformer  27   a  to  27   d  in  FIGS. 31 through 35 , may further include a first coupling winding wound to face the part of the first input winding  271  and a second coupling winding wound to face the second input winding  272 , and thus the second input winding  272  and the first input winding  271  are coupled through a distributed capacitance between the first input winding  271  and the first coupling winding and a distributed capacitance between the second input winding  272  and the second coupling winding, thereby cancelling out high frequency noise generated from the first input winding  271  and high frequency noise with the opposite polarity generated from the second input winding  272 . 
     [Fourth Embodiment] 
       FIG. 36  is an example of a configuration diagram illustrating a forward converter to which the transformer  19   a  of  FIG. 8  is applied. 
     The transformer  19   a  transfers energy through the input winding  191   a  and the output winding  193  with switching operation of the switching element  12  under the control of the driving circuit  18  using a voltage smoothened with the capacitor  11 . A negative output voltage is taken out through the output rectifier  14   a , the output rectifier  14   b , the inductor  29  and the capacitor  15 . Even in  FIG. 36 , a capacitive coupling between the input winding  191   a  and the output winding  193  is cancelled out with the cancellation winding  192   a , and as illustrated in  FIG. 9 , the number of turns of the cancellation winding  192   a  for cancellation may be set to be greater than that of the output winding  193 . 
       FIG. 37  is an example of a configuration diagram illustrating a forward converter to which the transformer  22   a  of  FIG. 18  is applied. 
     Referring to  FIG. 37 , the transformer  22   a  transfers energy through the first input winding  221   a , the second input winding  222   a , and the output winding  223  with switching operation of the switching element  12  under the control of the driving circuit  18 , and a capacitive coupling between the first input winding  221   a  and the output winding  223  is cancelled out by a capacitive coupling between the second input winding  222   a  and the output winding  223 , and the number of turns of the second input winding  222   a  for cancellation may be set to be greater than that of the output winding  223 . The other elements correspond to those of  FIG. 36 . 
     As described above, a flyback converter having a sandwich structure according to the present invention may have advantages such as generating far lower noise as well as transferring high efficient energy, low radiated noise due to high frequency noise cancellation, not requiring additional auxiliary windings for taking out auxiliary power sources, and having a simple structure of the transformer not requiring line filter reinforcement, thus greatly reducing the production cost.