Abstract:
A push-pull inverter is supplied from an inductively current-limited DC voltage source by way of a center-tap on a transformer having significant inductance. This transformer inductance is parallel-coupled with a capacitance means. The inverter is made to self-oscillate through positive feedback provided by way of a saturable current transformer. The inverter frequency is determined by the saturation time of this current transformer, which saturation time is designed to be somewhat longer than the half-cycle period of the natural resonance frequency of the transformer inductance combined with the capacitance means. By controlling the length of this saturation time, the magnitude of the current provided to the fluorescent lamp is controlled, thereby permitting control of the light output in response to changes in the magnitude of the power line voltage.

Description:
RELATED APPLICATIONS 
     The present application is a continuation of application Ser. No. 07/719,671 filed Jun. 24, 1991, now abandoned which is a continuation of application Ser. No. 07/346,292 filed May 1, 1989 now abandoned, which a continuation of application Ser. No. 06/697,949 filed Feb. 2, 1985, now abandoned; which was a Continuation-in-Part of Ser. No. 06/658,423 filed Oct. 4, 1984 now abandoned. 
    
    
     BACKGROUND OF THE INVENTION 
     1. Field of Invention 
     The present invention relates to power-line-operated inverter-type fluorescent lamp ballasts, particularly of the kind using a current-excited parallel-loaded resonant L-C circuit for matching the inverter output to the lamp load. 
     2. Prior Art 
     Power-line-operated inverter-type fluorescent lamp ballasts using a current-excited parallel-loaded resonant (or near resonant) L-C circuit for matching the inverter output to the lamp load are well known and widely used. An example of such a ballast is described in U.S. Pat. No. 4,277,726 to Burke. 
     One significant problem generally associated with these ballasts relates to their relatively poor ability to regulate lamp light output in response to variations in the magnitude of the power line voltage. 
     SUMMARY OF THE INVENTION 
     Objects of the Invention 
     An object of the present invention is that of providing cost-effective means for controlling and/or regulating the light output associated with inverter-type fluorescent lamp ballasts. 
     This as well as other important objects and advantages of the present invention will become apparent from the following description. 
     Brief Description 
     A push-pull inverter is supplied from an inductively current-limited DC voltage source by way of a center-tap on a transformer having significant inductance. This transformer inductance is parallel-coupled to a capacitance means. A fluorescent lamp is series-connected with a current-limiting inductor, and this lamp-inductor series-combination is connected in parallel with the capacitance means. 
     The inverter is made to self-oscillate through positive feedback provided by way of a saturable current transformer. The inverter frequency is determined by the saturation time of this current transformer, which saturation time is designed to be somewhat longer than the half-cycle period of the natural resonance frequency of the transformer inductance means combined with the parallel-coupled capacitance means as well as the parallel-coupled lamp-inductor series-combination. 
     The saturable current transformer comprises a ferrite magnetic core, and the length of the saturation time of this transformer is determined by the magnitude of the magnetic saturation flux of this ferrite core. 
     The magnitude of the magnetic saturation flux is determined by the temperature of the ferrite magnetic core: the higher the temperature, the lower the magnitude of the magnetic saturation flux. By heating the ferrite core as a function of the magnitude of the DC voltage supplying power to the inverter, the inverter frequency is made to vary in inverse relationship with the magnitude of this DC voltage. As an overall result, lamp light output is kept substantially constant in spite of substantial variations in the magnitude of the DC supply voltage. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 provides a schematic circuit diagram of the preferred embodiment of the invention. 
     FIG. 2 provides illustration of various voltage waveforms associated with the embodiment of FIG.  1 . 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENT 
     Details of Construction 
     FIG. 1 shows an AC voltage source S, which in reality is an ordinary 120 Volt/60 Hz electric utility power line. Connected directly across S is a bridge rectifier BR, the DC output from which is applied to a B+ terminal and a B− terminal—with the B+ terminal being of positive polarity in respect to the B− terminal. Connected directly between the B+ and B− terminals is an energy-storing capacitor ESC. 
     Connected between the B+ terminal and a center-tap PCT of the primary winding PW of a power transformer PT is an inductor means IM. Primary winding PW has two terminals PW 1  and PW 2  in addition to its center-tap PCT. Across terminals PW 1  and PW 2  is connected a capacitor C, as well as a series-combination of a fluorescent lamp FL and an inductor L. 
     Power transformer PT has two secondary windings SW 1  and SW 2  connected respectively with cathodes FLC 1  and FLC 2  of fluorescent lamp FL. 
     The collector of a first transistor Q 1  is connected to terminal PW 1  by way of a first primary winding CTp 1  of a saturable feedback current transformer CT; and the collector of a second transistor Q 2  is connected to terminal PW 2  by way of a second primary winding CTp 2  of current transformer CT. 
     Secondary winding CTs of current transformer CT is connected directly between the bases of transistors Q 1  and Q 2 . Current transformer CT has a ferrite magnetic core FMC. 
     A diode D 1  is connected between the base and the emitter of transistor Q 1 , with the diode&#39;s cathode being connected to the base. Similarly, a diode D 2  is connected between the base and the emitter of transistor Q 2 , with the diode&#39;s cathode being connected with the base. The emitters of transistors Q 1  and Q 2  are both connected with the B− terminal. Connected between the B+ terminal and the base of transistor Q 1  is a resistor R 1 . 
     A Zener diode Z is connected with its cathode to the B+ terminal and with its anode to one terminal of a resistor R 2 . The other terminal of resistor R 2  is connected with the B− terminal. 
     Resistor R 2  is placed in close proximity with the ferrite magnetic core FMC of current transformer CT; and a thermal conduction means TC is placed between resistor R 2  and this ferrite magnetic core FMC. 
     Details of Operation 
     In FIG. 1, the 120 Volt/60 Hz power line voltage is rectified by rectifier BR and, due to the filtering effect of capacitor ESC, provides a DC voltage of substantially constant magnitude between the B+ and the B− terminals. At low line voltage, the magnitude of this DC voltage is about 144 Volt; at normal line voltage it is 160 Volt; at high line voltage, it is 176 Volt. 
     The Zenering voltage of Zener diode Z is about 144 Volt; which implies that substantially no current flows through resistor R 2  at low line voltage. At high line voltage, on the other hand, current-flow through the Zener diode is only limited by the resistance of resistor R 2 , which at this point sees a voltage of about 32 Volt magnitude. The value of resistor R 2  is chosen such as to provide a certain amount of heating of the ferrite magnetic core of current transformer CT. 
     The purpose of providing heat to the ferrite magnetic core is that of controlling the oscillating frequency of the inverter, which oscillating frequency is principally determined by the saturation time associated with the saturable feedback current transformer CT. This saturation time is principally determined by the saturation magnetic flux density of the ferrite magnetic core FMC; and the saturation flux density, in turn, is a function of the temperature of the ferrite magnetic core: the higher the core temperature, the lower the magnitude of the saturation flux density. 
     Otherwise, the operation of the inverter circuit itself, which consists of inductor means IM, power transformer PT, saturable feedback current transformer CT, transistors Q 1  and Q 2 , diodes D 1  and D 2 , and biasing resistor R 1 , may be understood as a very special combination of the basic inverter circuits described in U.S. Pat. No. 4,277,726 to Burke and in U.S. Pat. No. 4,279,011 to Nilssen. By combining features of these two types of inverter circuits, useful effects may be obtained. 
     In the basic inverter circuit described by Burke, inverter oscillating frequency is determined by the inductance of power transformer PT as naturally interacting with the net capacitive reactance parallel-connected therewith (i.e., predominantly C). 
     On the other hand, in the basic inverter circuit described by Nilssen, inverter oscillating frequency is determined by the saturation time of the saturable feedback current transformer. 
     Thus, in the circuit of FIG. 1, inverter oscillating frequency is principally determined by the saturation time of saturable feedback current transformer PCT. However, for this to be the case, it is necessary that this saturation time be longer than the half-period of the natural resonance frequency associated with the inductance of the power transformer as interacting with the net capacitive reactance represented by capacitor C and the effect of the load circuit (which consists of inductor L in series-connection with the fluorescent lamp). 
     By way of various voltage waveforms, FIG. 2 effectively illustrates the operation of the inverter circuit of FIG.  1 . 
     FIG. 2 a  shows the intermittently sinusoidal waveform of the voltage present across power transformer PT as observed between its center-tap PCT and terminal PW 1  for the situation of low line voltage; in which situation the magnitude of the DC supply voltage is 144 Volt. Thus, in this situation, the average magnitude of the voltage existing during a complete half-cycle (i.e., between points Xa and Ya in FIG. 2 a ) must by basic necessity be 144 Volt. 
     By choice of saturation time of current transformer CT, and with no heating provided by resistor R 2 , the width Wa of this complete half-cycle is made to be about 22% wider than the width Ws of the base of the sinusoidal half-cycle existing between points Xa and Ya. As a necessary consequence of this fact, the amplitude of the sinusoidal half-cycle per se must be about 22% larger than would have been the case if the width Ws of this sinusoidal half-cycle had been equal to that of the complete inverter half-cycle Wa—the reason being that the average magnitude of the complete inverter half cycle must in this case by basic necessity be equal to 144 Volt. 
     In FIG. 2 a , it is noted that the sinusoidal half-cycle located between points Xa and Ya is a half-cycle of the natural resonance action between the inductance of power transformer PT and the combination of capacitor C and the load circuit consisting of inductor L connected in series with the fluorescent lamp FL. Thus, the degree that the distance between points Xa and Ya is longer than the base of the sinusoidal half-cycle corresponds to the degree that the inverter&#39;s oscillating frequency is lower than the natural resonance frequency of the inductance of power transformer PT as interacting with the combination of capacitor C and the load circuit. 
     FIG. 2 b  shows the corresponding voltage waveforms existing under the condition of normal line voltage, in which case the magnitude of the DC supply voltage is 160 Volt. In this case, due to heating by resistor R 2  of the ferrite magnetic core of transformer CT, the saturation time of this saturable transformer has been shortened to the point where the width Wb of the complete inverter half-cycle is only about 11% wider than the width Ws of the sinusoidal half-cycle. In this case, the average magnitude of the voltage existing between points Xb and Yb must by necessity be equal to 160 Volt; which implies that the magnitude of the sinusoidal half-cycle per se must be about 11% larger than it would have been if Ws had been equal to Wb. 
     FIG. 2 c  shows the corresponding voltage waveforms existing under the condition of maximum line voltage, in which case the magnitude of the DC supply voltage is 176 Volt. In this case, due to further heating by resistor R 2  of the ferrite magnetic core of transformer CT, the saturation time has been been shortened to the point where the width of Wc of the complete inverter half-cycle is about equal to the width Ws of the sinusoidal half-cycle. In this case, the average magnitude of the voltage existing between points Xc and Yc must by basic necessity be equal to 176 Volt. Also, at this point, the inverter oscillating frequency equals the natural resonance frequency of the inductance of power transformer PT as interacting with the combination of capacitor C and the load circuit. 
     Comparing FIGS. 2 a ,  2   b  and  2   c , it is seen that the magnitude of the sinusoidal half-cycles remains roughly constant even though the magnitude of the DC supply voltage varies over a range of +/−10%. The inverter frequency, on the other hand, changed substantially in inverse proportion to the magnitude of the DC supply voltage. With an inductor as the lamp current-limiting means, the overall implication is such that the power provided to the lamp remains substantially constant as the magnitude of the DC supply voltage changes by +/−10%. 
     The degree of controlling effect achieved depends on the degree of heating provided by resistor R 2  to the magnetic ferrite core of transformer CT: the lower the resistance of R 2 , the better the thermal coupling between R 2  and the magnetic ferrite core MFC, the smaller the physical size of transformer CT, the lower the heat losses from MFC, etc., the larger the degree of control. It is readily possible to provide for a change in the saturation time of some +/−20% as a result of the +/−10% change in the magnitude of the DC supply voltage, in which case the result would be over-regulation. 
     It is readily possible to provide for an additional and/or separate means for controlling the amount of heating provided to the ferrite core. For instance, a variable resistor means could be used to provide a selectable initial temperature bias; which, inter alia, could be used for control of light output irrespective of the magnitude of the DC supply voltage. 
     Or, the input from a light control means, such as a photo-cell, could be used for automatically affecting the core temperature and thereby to correspondingly control the amount of light provided by the lamp. 
     If, instead of an inductor, a capacitor were to be used as a lamp current-limiting means, it would still be possible to achieve light output regulation by way of controlling the saturation flux limits of the magnetic ferrite core of the saturable feedback transformer. However, in this case it would be necessary to heat the magnetic ferrite material in inverse relationship with the magnitude of the DC supply voltage. 
     By making R 2  non-linear (such as, for instance, combining it with a Zener diode and/or a thermistor), it is possible to provide an improved regulation profile, thereby attaining more accurate regulation of light output as function of line voltage magnitude variations. 
     The voltage waveforms illustrated by FIG. 2, such as the one shown by FIG. 2 a , correspond to the voltage provided to the series-combination of inductor L and fluorescent lamp FL. These waveforms may be described as sinusoidal half-cycles interconnected with periods of substantially zero voltage. 
     Also, by selectively modifying the functional relationship between the magnitude of the DC supply voltage and the amount of heat provided to the ferrite magnetic core, it is possible to arrange for the net inverter output voltage (as illustrated by FIG. 2) to behave in a wide variety of different ways: it would readily be possible to make the RMS magnitude remain constant as the magnitude of the DC supply voltage changes; or, to have the power provided to the load circuit (i.e., to the lamp) remain constant as the magnitude of the DC supply voltage changes; or, to have the magnitude of the fundamental frequency-component of the waveforms in FIG. 2 decrease in direct proportion with the basic repetition frequency of the waveforms (i.e., the basic inversion frequency); or, conversely, to make this repetition frequency decrease in direct proportion to the magnitude of this fundamental frequency component; etc. 
     In respect to the effect of temperature on the magnitude of the magnetic saturation flux of ferrite magnetic cores, reference is made to various handbooks and product catalogs relating to magnetic ferrites for inverter applications. 
     It is believed that the present invention and its several attendant advantages and features will be understood from the preceeding description. However, without departing from the spirit of the invention, changes may be made in its form and in the construction and interrelationships of its component parts, the form herein presented merely representing the presently preferred embodiment.