Abstract:
A submodule of a high-voltage inverter has a first sub-unit with a first energy storage device, a first series circuit of two power semiconductor switching units connected in parallel with the first energy storage device, each including a switchable power semiconductor, having the same pass-through direction, and each being conductive opposite the nominal pass-through direction. A first connection terminal is connected to the potential point between the power semiconductor switching units of the first series circuit. A second sub-unit has a second energy storage device, a second series circuit of two power semiconductor switching units connected in parallel with the second energy storage device, each including a switchable power semiconductor, having the same pass-through direction, and each being conductive opposite the nominal pass-through direction. A second connection terminal is connected to the potential point between the power semiconductor switching units of the second series circuit, limiting short circuit currents quickly, reliably, and effectively in case of a fault. The first and second sub-units are connected to each other by connections designed such that a current flow between the first connection terminal and the second connection terminal in both directions takes place only via the first energy storage device and/or the second energy storage device in a selected switching state of all power semiconductor switching units.

Description:
BACKGROUND OF THE INVENTION 
     Field of the Invention 
     The invention relates to a sub-module for forming a converter comprising a first sub-unit, which has a first energy storage device, a first series circuit—connected in parallel with the first energy storage device—formed by two power semiconductor switching units, which each have a power semiconductor that can be turned on and off and has the same forward direction, and are each conductive counter to said forward direction, and a first connection terminal, which is connected to the potential point between the power semiconductor switching units of the first series circuit, and a second sub-unit, which has a second energy storage device, a second series circuit—connected in parallel with the second energy storage device—formed by two power semiconductor switching units, which each have a power semiconductor that can be turned on and off and has the same forward direction, and are each conductive counter to said forward direction, and a second connection terminal, which is connected to the potential point between the power semiconductor switching units of the second series circuit. 
     The invention furthermore relates to a converter for example for high-voltage applications comprising power semiconductor valves, which in each case extend between an AC voltage connection and a DC voltage connection and form a bridge circuit, wherein each power semiconductor valve has a series circuit formed by two-pole sub-modules and each sub-module has at least one energy storage device and at least one power semiconductor circuit. 
     Such a sub-module and such a converter are already known from DE 101 03 031 A1. The converter described therein has power semiconductor valves connected to one another in a bridge circuit. Each of said power semiconductor valves has an AC voltage connection for connecting a phase of an AC voltage power supply system and a DC voltage connection, which can be connected to a pole of a DC voltage intermediate circuit. In this case, each power semiconductor valve consists of a series circuit formed by two-pole sub-modules each having a unipolar storage capacitor and a power semiconductor circuit connected in parallel with the storage capacitor. The power semiconductor circuit consists of a series circuit formed by power semiconductor switches, such as IGBTs or GTOs, for example, that can be turned on and off and are oriented in the same sense, with which power semiconductor switches a freewheeling diode is respectively connected in parallel in the opposite sense. One of two connection terminals of each sub-module is connected to the storage capacitor and the other connection terminal is connected to the potential point between the two power semiconductor switches that can be turned on and off. Depending on the switching state of the two drivable power semiconductor switches, either the capacitor voltage dropped across the storage capacitor or else a zero voltage can be applied to the two output terminals of the sub-module. On account of the series circuit formed by the sub-modules within the power semiconductor valve, a so-called multistage converter that impresses DC voltage is provided, wherein the level of the voltage steps is defined by the level of the respective capacitor voltage. Multistage or multipoint converters have the advantage over the two- or three-stage converters comprising central capacitor banks that high discharge currents are avoided upon a short circuit on the DC voltage side of the converter. Furthermore, in the case of multistage converters the outlay when filtering harmonics is reduced by comparison with two- or three-point converters. 
     Multipoint converters are also preferably suitable for constructing spatially extensive branched DC voltage power supply systems that are required particularly in the case of so-called off-shore wind farms and in connection with solar power networks in desert regions. 
     One important prerequisite for use of the converters in these fields is, however, reliable control of short circuits in the DC voltage power supply system. Expedient mechanical switches for extremely high DC voltages, which can switch high fault currents under load, are not available owing to fundamental physical problems. The technically achievable turn-off times and the switching overvoltage of mechanical switches are also troublesome. 
     EP 0 867 998 B1 describes the use of electronic power semiconductor switches in the DC voltage intermediate circuit of a high-voltage direct-current transmission system. However, the use of power semiconductor switches with DC voltages of a few hundred kilovolts is beset by the disadvantage that the high voltage necessitates a high number of power semiconductors connected in series. This also means, however, that a high on-state loss is also established at these components. Furthermore, it is necessary to provide overvoltage limiters connected in parallel with the power semiconductors, as a result of which the outlay is additionally increased. The overvoltage limiters generally do not have ideal limiter characteristic curves, and so the number of power semiconductors connected in series has to be designed to be even higher than would actually be required by the rated voltage. As a result of this overdimensioning, the on-state losses rise even further. 
     WO 2008/067786 A1 describes a multistage converter comprising series circuits formed by sub-modules, wherein each sub-module has a thyristor alongside a capacitor connected in parallel with a power semiconductor circuit. The thyristor is connected in parallel with a freewheeling diode of the power semiconductor circuit, which carries the entire short-circuit current in the case of a fault. The parallel thyristor is triggered in the case of a short circuit, thereby relieving the load on the freewheeling diode. 
     Alongside the abovementioned applications in the field of electrical energy transmission and distribution, multipoint converters that impress DC voltage are, of course, also outstandingly suitable for use in the field of drive technology. 
     The multipoint or multistage converters mentioned in the introduction have the disadvantage that a short-circuit current via the converter cannot be limited in both directions without additional measures, and so the semiconductors of the converter and external components in the shorted circuit are jeopardized or destroyed. 
     BRIEF SUMMARY OF THE INVENTION 
     It is an object of the invention to provide a sub-module and a converter of the type mentioned in the introduction with which short-circuit currents occurring in the case of a fault can be effectively limited and installation damage can be reliably avoided. Furthermore, faulty sections of a DC voltage power supply system are intended to be able to be de-energized as rapidly as possible and in this way isolated from the rest of the DC voltage power supply system. 
     Finally, in the case of a short circuit on the DC voltage side of the converter, the currents on the AC voltage side thereof are intended to be influenced as little as possible and triggering of the mechanical switches on the AC side is intended to be avoided. 
     Proceeding from the sub-module mentioned in the introduction, the invention achieves this object by virtue of the fact that the first sub-unit and the second sub-unit are connected to one another via connecting means configured such that in at least one selected switching state of all the power semiconductor switching units a current flow takes place between the first connection terminal and the second connection terminal in both directions only via the first energy storage device and/or the second energy storage device. 
     Proceeding from the converter mentioned in the introduction, the invention achieves this object by virtue of the fact that the sub-module is a sub-module according to the invention. 
     According to the invention, two sub-units each having an energy storage device, for example a capacitor, and a series circuit formed by two power semiconductor switching units are connected to one another via connecting means. The connecting means are embodied in a manner deviating from the prior art such that upon suitable driving of the power semiconductor switching units a current flow between the two connection terminals of the sub-module according to the invention always has to take place via at least one energy storage device. The energy storage device respectively affected always builds up, independently of the polarization of the clamping current, a back EMF that allows the current flow to decay rapidly. According to the invention, the selected switching state is dependent on the topology of the connecting means and the components thereof. 
     According to the invention, a high short-circuit current can be controlled without external additional switches. In contrast to the prior art, in the context of the invention it is ensured that high short-circuit currents through the converter itself in both directions can be avoided rapidly, reliably and effectively. Additional switches, for example in the DC voltage circuit connected to the converter, or else semiconductor switches connected in parallel with a power semiconductor of the sub-module, have become superfluous in the context of the invention. In the case of a fault, the sub-modules according to the invention virtually exclusively take up the liberated energy, such that the latter is completely absorbed. The energy absorption results in a back EMF and can be dimensioned in a defined and desired manner, such that unfavorably high voltages are avoided. Furthermore, according to the invention, no energy storage devices have to be charged in a controlled manner for restarting the converter. Rather, the converter according to the invention can resume its normal operation at any time. 
     Expediently, the connecting means have a switching unit. Said switching unit is in its interruption position, for example, in said selected state. In a departure therefrom, however, it is also possible according to the invention for the switching unit to be in its continuity position in the selected switching state. The design of the switching unit is arbitrary, in principle, in the context of this further development of the invention. Thus, it can be, for example, a mechanical switching unit, a suitable semiconductor switch, or else a power semiconductor switching unit that is identical to the rest of the power semiconductor units of the converter. The configuration of the power semiconductor switching units will be discussed in even more detail later. 
     Expediently, the connecting means have at least one potential isolating diode which is designed to maintain a voltage difference between the first sub-unit and the second sub-unit. In accordance with this advantageous further development, it is possible to increase the number of voltage steps that can be achieved. Thus, it is possible, for example, to generate the sum of the voltages dropped across the first energy storage device and across the second energy storage device at the connection terminals of the sub-module. Furthermore, in the case of this configuration of the invention, there is the possibility of generating, depending on the switching state of the power semiconductor switching units, only one voltage, that is to say either the voltage dropped across the first energy storage device or the voltage dropped across the second energy storage device, at connection terminals. In this way, the first and second sub-units can be treated, from the standpoint of control engineering, like two sub-modules in accordance with the prior art. Previously established control methods can thus also be applied to the sub-module according to the invention. 
     Furthermore, it is advantageous for the connecting means to have at least one damping resistor. The damping resistor or resistors support(s) the energy storage devices in taking up energy in the case of a fault. For this purpose, the damping resistors are interconnected with the remaining components of the connecting means in such a way that in said selected switching state a current flow passes at least partly also via the damping resistors independently of the polarity of the clamping current. 
     In accordance with one preferred configuration of the invention, the connecting means have an emitter connecting branch, which connects an emitter of a first power semiconductor switching unit of the first series circuit to an emitter of a first power semiconductor switching unit of the second series circuit and in which a potential isolating diode is arranged, a connector collecting branch, which connects a collector of the second power semiconductor switching unit of the first series circuit to a collector of the second power semiconductor switching unit of the second series circuit and in which a potential isolating diode is arranged, and a switching branch, in which a switching unit is arranged and which connects the cathode of the potential isolating diode of the emitter connecting branch to the anode of the potential isolating diode of the collector connecting branch. The emitter of a power semiconductor switching unit is also designated as source or cathode. 
     In accordance with one expedient further development in this regard, a respective damping resistor is arranged in the emitter connecting branch and in the collector connecting branch. As has already been explained, the switching unit of the switching branch can be chosen arbitrarily, in principle. What is essential is that the switching unit can be switched back and forth between an interrupter position, in which it interrupts a current flow, and an on-state position, in which it is conducting. It is thus possible, for example, to use as switching unit a mechanical circuit breaker, a cost-effective semiconductor switch or else a power semiconductor switching unit that is identical to the rest of the power semiconductor switching units of the sub-module. Other drivable power semiconductors can also be used as switching unit in the context of the invention. 
     As has already been explained, in accordance with this expedient further development, the selected switching state is attained if all the power semiconductor switching units and the switching unit are in their interrupter position. The clamping current is then always passed via at least one energy storage device or a damping resistor. 
     The switching unit should always be chosen such that the power loss arising at it during normal operation of the sub-module is as low as possible. 
     If all the power semiconductor switching units of the sub-module are designed identically, in other words if all the semiconductor switches are identical, they have a uniform reverse voltage and structure. This is advantageous in the case of high voltages, because only few semiconductor switches are suitable for extremely high voltages and powers. Uniform equipment of the sub-modules makes it possible to use the best suited and most powerful semiconductors in each case. 
     Expediently, each power semiconductor switching unit has a respective power semiconductor that can be turned on and off, with which power semiconductor a freewheeling diode is connected in parallel in the opposite sense. Such power semiconductors that can be turned off are, for example, commercially available IGBTs or GTOs and the like. These power semiconductors are usually used with a freewheeling diode connected in parallel in the opposite sense. However, reverse conducting power semiconductors can also be used according to the invention. Separate freewheeling diodes are then unnecessary. 
     Expediently, each energy storage device is configured as a capacitor, but in particular as a unipolar storage capacitor. 
     Further advantages and configurations are the subject matter of the following description of exemplary embodiments with reference to accompanying figures of the drawing, wherein identical reference signs refer to identically acting component parts and wherein 
    
    
     
       BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWING 
         FIG. 1  schematically illustrates an exemplary embodiment of the converter according to the invention, and 
         FIG. 2  illustrates in greater detail an exemplary embodiment of the sub-modules according to the invention. 
     
    
    
     DESCRIPTION OF THE INVENTION 
       FIG. 1  shows an exemplary embodiment of the converter  1  according to the invention in a schematic illustration. It can be discerned that the converter  1  has power semiconductor valves  2  connected to one another in a bridge circuit. Each of the power semiconductor valves  2  extends between an AC voltage connection L 1 , L 2 , L 3  and a DC voltage connection  3   1 ,  3   2 ,  3   3  and  4   1 ,  4   2 ,  4   3 , respectively. The DC voltage connections  3   1 ,  3   2 ,  3   3  can be connected via a positive pole connection  5  to a positive pole and via a negative pole connection  6  to a negative pole of a DC voltage power supply system, not illustrated in the figures. 
     The AC voltage connections L 1 , L 2  and L 3  are in each case connected to a secondary winding of a transformer, the primary winding of which is connected to an AC voltage power supply system, likewise not illustrated in the figures. An AC voltage connection L 1 , L 2 , L 3  is provided for each phase of the AC voltage power supply system. In the exemplary embodiment shown, the AC voltage power supply system is a three-phase system. Consequently, the converter  1  also has three AC voltage connections L 1 , L 2 , L 3 . Mechanical circuit breakers are expediently provided between the AC voltage connection L 1 , L 2 , L 3  and the transformer in order to isolate the AC voltage power supply system from the converter  1  in the case of a fault. The circuit breakers are likewise not illustrated in  FIG. 1 . 
     In the exemplary embodiment shown, the converter  1  is part of a high-voltage direct-current transmission installation and serves for connecting AC voltage power supply systems in order to transmit high electrical powers between them. It should be mentioned at this juncture, however, that the converter can also be part of a so-called FACTS installation serving for system stabilization or serving to ensure a desired voltage quality. Furthermore, it is also possible to use the converter in accordance with  FIGS. 1 and 2  in drive technology. 
     In  FIG. 1  it can furthermore be discerned that each power semiconductor valve  2  has a series circuit composed of sub-modules  7  and also an inductor  8 . In this case, each sub-module  7  has two connection terminals x 1  and x 2 . 
       FIG. 2  shows an exemplary embodiment of the sub-module  7  according to the invention in greater detail. It should be pointed out at this juncture that all of the sub-modules  7  illustrated schematically in  FIG. 1  are constructed identically.  FIG. 2  therefore shows the construction of all the sub-modules  7  and the converter  1  in representative fashion on the basis of one sub-module  7 . 
     The sub-module  7  in accordance with  FIG. 2  has a first sub-unit  9  and a second sub-unit  10 , which are framed by a dashed line and are constructed identically. Thus, the first sub-unit  9  comprises a first series circuit  11  composed of power semiconductor switching units  12  and  13 , which, in the exemplary embodiment shown, have a respective IGBT  14  and  15  as power semiconductor that can be turned on and off, and a respective freewheeling diode  16  and  17 , which is connected in parallel with the respectively assigned IGBT  14 ,  15  in the opposite sense. The IGBTs  14 ,  15  have the same forward direction, that is to say are oriented in the same sense. The potential point between the power semiconductor switching units  12  and  13  is connected to a first connection terminal x 2 . The series circuit  11  is connected in parallel with the first capacitor  18  as first energy storage device, across which the voltage U C1  is dropped. 
     The second sub-unit  10  comprises a second series circuit  19  comprised of a first power semiconductor switching unit  20  and a second power semiconductor switching unit  21 , which have a respective IGBT  22  and  23  as power semiconductor that can be turned on and off. The IGBTs  22 ,  23  have the same forward direction in the series circuit  19 , such that the power semiconductor switching units  20  and  21  are oriented in the same sense. A freewheeling diode  24  and  25  is connected in parallel with each IGBT  22  and  23 , respectively, of the second series circuit  19  in the opposite sense. The second series circuit  19  is connected in parallel with a second capacitor  26 , across which the voltage U C2  is dropped. The potential point between the power semiconductor switching units  20  and  21  is connected to the second connection terminal x 1 . 
     The sub-units  9  and  10  are linked to one another via connecting means  27 . The connecting means  27  have an emitter connecting branch  28  and also a collector connecting branch  29 . The emitter connecting branch  28  connects the emitter of the IGBT  15  of the first series circuit  11  to the emitter of the IGBT  23  of the second series circuit  19 . The collector connecting branch  29 , by contrast, connects the collector of the IGBT  14  of the first series circuit  11  to the collector of the IGBT  22  of the second series circuit  19 . A potential isolating diode  30  and a limiting resistor  31  are arranged in the emitter connecting branch  28 . The collector connecting branch  29  likewise has a potential isolating diode  32  and also a limiting resistor  33 . The emitter connecting branch  28  is connected to the collector connecting branch  29  via a switching branch  34 , in which a switching unit  35  is arranged. In the exemplary embodiment shown, the switching unit is realized as a power semiconductor switching unit  35  and comprises an IGBT  36  and a freewheeling diode  37  connected in parallel therewith in the opposite sense. In this case, the switching branch  34  connects the cathode side of the potential isolating diode  30  to the anode side of the potential isolating diode  32 , the limiting resistor  33  arranged between said anode and the switching branch  34  having been disregarded. 
     The mode of operation of the circuit of the sub-modules  7  is explained below. Firstly, it should be pointed out that the required reverse voltage of all the power semiconductors, that is to say both of the freewheeling diodes  16 ,  17 ,  24  and  25  and of the power semiconductor switches  14 ,  15 ,  22  and  23  that can be turned on and off, depends on the maximum voltage of the two unipolar storage capacitors  18  and  26 , which is identical in the exemplary embodiment chosen. A disadvantageous overdimensioning of the reverse voltages of said power semiconductors is avoided in this way. 
     Overall it is possible to differentiate between a plurality of switching states that differ from one another with regard to the clamping voltages U x . 
     In one switching state  1  picked out by way of example, the clamping voltage U x  dropped across the connection terminals, x 2  and x 1  is equal to zero independently of the direction of the clamping current. In this switching state, the IGBTs  15 ,  22  and  36  are situated in their on-state position, in which a current flow is made possible in the forward direction via the respective IGBT. The remaining IGBTs, that is to say the IGBTs  14  and  23 , by contrast, are situated in their blocking position, such that a current flow via said IGBTs is interrupted. Given a positive current direction i x  (i x  positive), which is indicated by the arrow at the first connection terminal x 2  in  FIG. 2 , the power semiconductors  15 ,  37  and  22  are current-carrying. Given a negative current direction (i x  negative), the power semiconductors  24 ,  36  and  17  are current-carrying. 
     The following table summarizes the switching states preferably used. 
     
       
         
               
               
               
               
               
               
               
               
               
               
             
               
               
               
               
               
               
               
               
               
               
             
           
               
                   
               
               
                 Switching 
                   
                 IGBT 
                 IGBT 
                 IGBT 
                 IGBT 
                 IGBT 
                   
                   
                   
               
               
                 state 
                 i x   
                 15 
                 14 
                 23 
                 22 
                 36 
                 U X   
                 W C1   
                 W C2   
               
               
                   
               
             
             
               
                   
               
             
          
           
               
                 1 
                 negative 
                 1 
                 0 
                 0 
                 1 
                 1 
                 0 
                 0 
                 0 
               
               
                   
                 positive 
                 1 
                 0 
                 0 
                 1 
                 1 
                 0 
                 0 
                 0 
               
               
                 2 
                 negative 
                 0 
                 1 
                 0 
                 1 
                 1 
                 +U C1   
                 −1 
                 0 
               
               
                   
                 positive 
                 0 
                 1 
                 0 
                 1 
                 1 
                 +U C1   
                 +1 
                 0 
               
               
                 3 
                 negative 
                 0 
                 1 
                 1 
                 0 
                 1 
                 +(U C1  + U C2 ) 
                 −1 
                 −1 
               
               
                   
                 positive 
                 0 
                 1 
                 1 
                 0 
                 1 
                 +(U C1  + U C2 ) 
                 +1 
                 +1 
               
               
                 4 
                 negative 
                 1 
                 0 
                 1 
                 0 
                 1 
                 +U C2   
                 0 
                 −1 
               
               
                   
                 positive 
                 1 
                 0 
                 1 
                 0 
                 1 
                 +U C2   
                 0 
                 +1 
               
               
                 5 
                 negative 
                 0 
                 0 
                 0 
                 0 
                 0 
                   −(U C1  + U C2 )/2 
                 +1 
                 +1 
               
               
                   
                 positive 
                 0 
                 0 
                 0 
                 0 
                 0 
                 +(U C1  + U C2 ) 
                 +1 
                 +1 
               
               
                   
               
             
          
         
       
     
     The columns W C1  and W C2  are intended to illustrate whether the storage capacitors  18  and  26  take up or output energy, where +1 stands for taking up energy and −1 stands for outputting energy. 
     It can be gathered from the table that a positive voltage is always generated at the connection terminals x 2  and x 1  in the switching states  2 ,  3  and  4 . This holds true independently of the direction of the clamping current. Thus, by way of example, the capacitor voltage U C1  or the capacitor voltage U C2  or else the sum of the capacitor voltages U C1 +U C2  can be dropped across the connection terminals. 
     In the switching state  5 , all the drivable power semiconductors, that is to say the IGBTs  14 ,  15 ,  22 ,  23  and  36 , are in their interrupter position, such that a current flow via the IGBTs is interrupted. In this switching state, the clamping voltage U x  always forms a back EMF independently of the polarity of the clamping current i x , such that the sub-module  7  always absorbs energy. Given a negative current direction, i x  negative, a negative back EMF is generated by the parallel circuit formed by the storage capacitors  26  and  18  and also by the voltage drop across the damping resistors  30  and  32 . If the capacitor voltages U C1  and U C2  do not correspond exactly, they are automatically balanced. In the switching state  5 , the following holds true to a good approximation: 
               U   x     =         -     (       U     c   1       +     U     c   2         )       2     -     U   R             
where U R  corresponds to the voltage drop across the damping resistors  32  and  30 .
 
     Given a positive current direction, a positive back EMF
 
 U   x =+( U   C1   +U   C2 )
 
is generated. Here, too, a current flow can take place only with the charging of the storage capacitors  18  and  25 , respectively. In this case, it is advantageous that the current that occurs is passed via both capacitors, since then there occurs at the latter a lower overvoltage than if only one capacitor had to take up the energy.
 
     It can furthermore be gathered from the table presented above that with the sub-module  7  and its two sub-units  9  and  10  it is possible to generate the same output voltages at the output terminals as in the case of two series-connected sub-modules in accordance with the prior art (DE 101 03 031 A1). The sub-units  9 ,  10  correspond as it were to a respective sub-module in accordance with the prior art. In other words, the sub-module according to the invention in accordance with  FIG. 2  can be driven in the same way as two sub-modules in accordance with the prior art. All known control methods can therefore still be employed. In the narrower sense, however, the secondary condition exists that the number of sub-modules connected in series in accordance with the prior art must always yield an even number. In the case of high-voltage applications, however, the number of sub-modules connected in series is so large that said secondary condition is unimportant. 
     The switching state  5  can be used for complete current reduction in the case of a fault. If all sub-modules  7  are converted to this switching state, the branch currents of the converter  1  and in a resulting manner also the currents on the AC voltage side and DC voltage side, on account of the sum of the back EMFs of all the series-connected sub-modules  7 , are reduced very rapidly to zero. The speed of this current reduction results from the abovementioned back EMF and the inductances present in total in the electric circuits. It is typically of the order of magnitude of a few milliseconds in the exemplary embodiment shown. 
     The dead time until the beginning of the current reduction is substantially dependent on the response time of the switching unit  35 . If a power semiconductor switching unit in accordance with  FIG. 2  is used for the switching unit  35 , said dead time is negligible. The dead time is then substantially owing to the inertia of the various measuring sensors and current conductors used to identify a disturbance case. This inertia of this measured value detection is at the present time typically in the range of a few tens of microseconds. 
     The advantages of the sub-module according to the invention and of the converter  1  according to the invention can be summarized as follows: Firstly, the time period until complete reduction of a short-circuit current which occurs in the case of a fault is very short. Consequently, switches provided on the AC voltage side of the converter  1  do not even have to be triggered in the first place. Both the current on the AC voltage side and the current on the DC voltage side exceed the rated current only insignificantly. In contrast to the prior art, the power semiconductors of the sub-modules do not have to be protected by thyristors or other bridging elements. The reliability of the current turn-off is very high, because a redundancy is ensured by the large number of series-connected sub-modules in the power semiconductor valves of the converter  1 . In connection with reliability, it should also be explained that the converter  1  is in operation continuously with all its components and is constantly monitored metrologically. Such functional reliability is not afforded in comparable devices for current reduction in cases of fault, which are activated only in such a case of fault. 
     A further essential advantage of the invention is that “switching back” to normal operation is possible at any time, such that, even in the case of faulty unnecessary triggering or detection, the negative effects on installation operation can be minimized. 
     With the aid of a converter  1  according to the invention it is furthermore possible, even in a branched DC voltage power supply system, to bring the DC voltage currents rapidly to zero. In the DC voltage circuit, isolation without current, for example by means of vacuum interrupters or antiparallel thyristors, is possible in this way. In the case of branched DC voltage power supply systems, it is also necessary, of course, for the remaining converters connected to the DC voltage power supply system to reduce the current, that is to say to undergo transition rapidly into the switching state  5  of the sub-modules  7 . A faulty system section of the DC voltage power supply system can thus be isolated simply and cost-effectively from the rest of the DC voltage power supply system without current by means of known mechanical switches. The faulty system section can then “pause” for the purpose of deionization or fault localization and later be started up by its assigned converter. In a very short time, the remaining converters can reactivate the entire DC voltage power supply system.