Abstract:
The system includes a current injection device in electrical communication with the switch mode power supply. The current injection device is positioned to alter the initial, non-zero load current when activated. A prognostic control is in communication with the current injection device, controlling activation of the current injection device. A frequency detector is positioned to receive an output signal from the switch mode power supply and is able to count cycles in a sinusoidal wave within the output signal. An output device is in communication with the frequency detector. The output device outputs a result of the counted cycles, which are indicative of damage to an a remaining useful life of the switch mode power supply.

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This application claims priority to co-pending U.S. Provisional Application entitled “Health Monitoring in Switch-Mode Power Supplies with Voltage Regulation” having Ser. No. 60/831,310 filed Jul. 17, 2006, which is entirely incorporated herein by reference. 
    
    
     STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT 
     This invention was made in part with Government support under contract NNA06AA22C awarded by NASA Ames Research Center. The Government may have certain rights in this invention. 
    
    
     FIELD OF THE INVENTION 
     The present invention is generally related to detecting fault-to-failure signatures as precursors to failure of electronic devices; and is particularly related to detecting fault-to-failure signatures as precursors to failure of an optical isolator. 
     BACKGROUND OF THE INVENTION 
     Prognostic methods are used to improve the reliability of deployed systems by looking at components that have high failure rates and critical impact on performance within the systems. Detectors, or sensors, monitor these systems and look for failure precursors that indicate the high-failure rate components have entered a wear-out mode and are degrading toward failure. By knowing the progression of failure dynamics for a device, an accurate prediction of time to failure or remaining useful life (RUL) can be made and an appropriate maintenance action, such as remove and replace the device, can be initiated to avoid system failure during a time of operation. Fault-to-failure signature detection is a method or capability to detect and report a precursor-to-failure or incipient fault condition of a component device or assembly containing the component. Such detection is the basis for a notification capability to provide early warning of degradation and eventual failure. 
     A direct approach is to place sensors at the board level at each node of each component having a significant rate of failure: faults are detected and tracked. In many cases, there are interdependencies in these tracking measurements that require an expert system to produce RUL estimates with greater accuracy. This direct approach is invasive because it requires internal access to components within the system: adding sensors inside the power supply imposes an additional reliability load. Manufacturers of switch-mode power supplies can be reluctant to enable prognostics on their supplies, believing that the benefits of this capability do not justify the cost and/or that these benefits do not outweigh the additional reliability burden. Currently, this direct approach to prognostics has not been adopted in many applications. A non-invasive approach uses external access methods, such as using an output voltage terminal, to attach electronic equipment to measure values, inject stimuli and sense responses. 
       FIG. 1  is a simplified block diagram of a switch mode power supply, (hereinafter “SMPS”)  1 , in accordance with the known prior art. The SMPS  1  has a direct voltage input  3 . A SMPS uses relatively high-frequency switching devices, such as a power Metallic Oxide Semiconductor Field Effect Transistor (hereinafter “MOSFET”) switch, sometimes contained within a circuit. In  FIG. 1 , the MOSFET switch is contained within a Pulse Width Modulator (hereinafter “PWM”)  5 . Switching frequencies of one-hundred-thousand Hertz (100 kHz) or higher are used to convert the input direct voltage to a first pulsed waveform  7 . One or more high-frequency transformers in Isolation device  9  with one or more outputs are used to provide an isolation barrier and to buck or boost the first pulsed waveform  7  to a second pulsed waveform  11  having a different voltage amplitude. The second pulsed waveform  11  is rectified and filtered by an output filter  13  to produce one or more direct voltages with a positive terminal  15 A and a negative terminal  15 B at voltage levels different from the direct voltage input  3 . Feedback is provided through the use of either a small pulse transformer or an opto-isolator in a feedback circuit  17 . The feedback circuit  17  produces feedback output  19  to PWM  5  to control the pulse width or pulse frequency or both of the switching devices in the PWM  5  to regulate the output direct voltage across the terminals  15 A and  15 B. 
       FIG. 2  is a schematic diagram of an opto-isolator  21  from the feedback circuit  17  of  FIG. 1 , in accordance with the known prior art. Referring to  FIG. 1  and to  FIG. 2 , in an SMPS  1 , there are three components that are at the root of the majority of failures; (1) an output capacitor in the output filter  13 , (2) a power MOSFET switch in either or both the Pulse Width Modulator  5  and the Isolation device  9 , and (3) an opto-isolator in the feedback circuit  17 . The basic operation of an opto-isolator  21 , as shown in  FIG. 2 , is the following: an input current flows into a first opto-isolator input terminal  23 A, through a light-emitting diode  25  and out through a second opto-isolator input terminal  23 B; the current flow through the light-emitting diode  25  produces light  27 ; the produced light  27  causes current to flow in the base of an opto-isolator transistor  29 , which causes current to flow through the opto-isolator transistor  29  via output terminals  31 A and  31 B. The input current to the opto-isolator  21  is the result of circuitry to which the opto-isolator  21  is connected in feedback circuit  17  in  FIG. 1 ; the collector at the top of the opto-isolator transistor  29  is connected to circuitry in the Pulse Width Modulator  5  shown in  FIG. 1 . 
     The ratio of the output current to the input current of an opto-isolator  21  is defined as the Current Transfer Ratio (CTR) of the opto-isolator  21 . Feedback circuits  17  are designed to cause a regulated and set direct output voltage, V DC , across output terminals  15 A and  15 B in  FIG. 1 . The design of the feedback circuit  17  is predicated on a CTR greater than 1. As an opto-isolator  21  is stressed during operation, for example by heat or current, the crystal lattice of the semiconductor material comprising the opt-isolator  21  may develop defects, especially point defects. Thus, the opto-isolator  21  is damaged. Such defects reduce the light emitting efficiency of the light-emitting diode  25 , and the amplification factor of the opto-isolator transistor  27  is reduced, which reduces the CTR. When the CTR is reduced, the opto-isolator  21  is said to be operating in degraded state. When the stress conditions are removed, the crystal lattice self-anneals and most, but not all, of the lattice damage is repaired. After repeated cycles, the CTR is gradually reduced so low that the SMPS  1  fails to correctly regulate the direct output voltage, V DC , and the SMPS  1  is deemed to be defective. A defective SMPS  1  is typically removed and replaced, with the removed SMPS  1  being set aside for subsequent testing, evaluation and repair. Because it is set aside, the stress conditions are removed, the opto-isolator  21  self-anneals and the CTR increases. Often the increase in the CTR is sufficient to result in correct voltage regulation during a re-test, and the SMPS  1  is evaluated as not being defective. This intermittent fault behavior, defective in operation but okay during re-test, is a major contributor to a high number of SMPS&#39;s being swapped out as defective and being returned to regular usage after maintenance fails to duplicate the defective behavior. 
     The primary fault-to-failure progression of an opto-isolator  21  is a gradual decrease over time of the current transfer ratio (CTR) from a value greater than 1.0 and typically less than 3.0 to a value below 0.2. As the CTR decreases, the ability of the SMPS  1  to regulate the direct output voltage V DC  degrades until the SMPS  1  is unable to adequately regulate the direct output voltage V DC . Of interest are the following: (1) the identification and characterization of a fault-to-failure progression signature for opto-isolators  21  in feedback loops of SMPS&#39;s  1 ; (2) a method to use the fault-to-failure progression signature in the design and implementation of a non-invasive sensor for prognostication of opto-isolators  21  in the feedback loop; and (3) a measure of time remaining before the CTR will decrease to a value so low as to cause the power supply to fail to regulate the direct output voltage V DC . 
     There are a number of prognostic methods to identify failure precursors in switching power supplies that rely on internal or external measurements. One known method is to monitor ripple voltage at the output terminals of the power supply as discussed in Layyani, “Failure Prediction of Electrolytic Capacitors of a Switch Mode Power Supply, IEEE Transactions on Power Electronics, Vol 13, No. 6, November 1998. The precursor to failure of the method of Layyani is an increase in ripple voltage caused by increasing degradation of the capacitor as it fails. 
     A second method disclosed in U.S. Pat. No. 4,245,289, to Mineck, is to measure the duty cycle modulated by an integrated circuit (IC) component that is responsible for switch timing in a regulated power supply. Mineck is based on the premise that electronic components consume more power as they begin to fail. Overall efficiency decreases with the increased power consumption. As the output is regulated to produce a predetermined and set value, the switching duty cycle must compensate by changing the relative on- and off-times. The precursor-to-failure utilized in Mineck is an increase in duty cycle. The method of Mineck is non-specific with regard to exactly which component is failing. Furthermore, the method of Mineck is invasive because it requires measurement of an internal node voltage waveform. 
     An obvious prognostic method might be to measure the difference between the actual direct output voltage and a designed-for direct output voltage set point. Any trend in the difference value indicates that the power supply is degrading along a trajectory toward failure; but as a prognostic or precursor to failure for an opto-isolator  21 , this approach is limited because the SMPS  1  will continue to regulate the direct output voltage up to the point where the opto-isolator  21  actually fails: there is an absence of any precursor to failure from the direct output voltage. 
     None of the discussed prior art methods have the ability to detect increasing degradation in the operation of an opto-isolator  21  in a feedback loop of a SMPS  1 , and there is no known method or means to predict the failure of this component using only measurement at external nodes, such as the direct output voltage terminal of the SMPS  1 . 
     There are existing patents concerned with monitoring of the direct output voltage and feedback circuits  17 . In those cases, the monitoring is used for control and provides no information regarding the health of the feedback components. These inventions are in a different category from the prognostic inventions. There are also patent applications related to processing of monitored data in switching supplies, such as US patent application publication no. 20050289378 in which an integrated circuit approach for multi-parameter monitoring is proposed. Patent application publication no. 20030039129 detects an abrupt load change and is implemented as part of a strategy to reduce overshoot and improve overall transient response. 
     SUMMARY OF THE INVENTION 
     Embodiments of the present invention provide a system and method for estimating a remaining useful life of a switch mode power supply having an initial, non-zero load current. Briefly described, in architecture, one embodiment of the system, among others, can be implemented as follows. The system includes a current injection device in electrical communication with the switch mode power supply. The current injection device is positioned to alter the initial, non-zero load current when activated. A prognostic control is in communication with the current injection device, controlling activation of the current injection device. A frequency detector is positioned to receive an output signal from the switch mode power supply and is able to count cycles in a sinusoidal wave within the output signal. An output device is in communication with the frequency detector. The output device outputs a result of the counted cycles. 
     The present invention can also be viewed as providing methods for estimating a remaining useful life of a switch mode power supply having an initial, non-zero load current. In this regard, one embodiment of such a method, among others, can be broadly summarized by the following steps: detecting a damped ringing response to a change in the load current; counting a plurality of cycles in the damped ringing response; and outputting the plurality of cycles count, whereby the plurality of cycles count is indicative of damage to the switch mode power supply and at least one opto-isolator therein. 
     Other systems, methods, features, and advantages of the present invention will be or become apparent to one with skill in the art upon examination of the following drawings and detailed description. It is intended that all such additional systems, methods, features, and advantages be included within this description, be within the scope of the present invention, and be protected by the accompanying claims. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Many aspects of the invention can be better understood with reference to the following drawings. The components in the drawings are not necessarily to scale, emphasis instead being placed upon clearly illustrating the principles of the present invention. Moreover, in the drawings, like reference numerals designate corresponding parts throughout the several views. 
         FIG. 1  is a simplified block diagram of a switch mode power supply, in accordance with the prior art. 
         FIG. 2  is a schematic diagram of an opto-isolator from the feedback circuit of  FIG. 1 , in accordance with the known prior art. 
         FIG. 3  is a line plot showing a ringing response, such as that in response to an abrupt change in current, on a 5.0 V output voltage of a SMPS. 
         FIG. 4  is a line plot of the gain (A) in dB and the phase (B) in degrees of a SMPS versus frequency. 
         FIG. 5  is a schematic diagram of a first exemplary embodiment of a feedback circuit for a SMPS. 
         FIG. 6  is a schematic diagram of an exemplary current injection device, in accordance with a first exemplary embodiment of the present invention. 
         FIG. 7  is a block diagram of a SMPS health monitor connected to the output voltage bus of a SMPS, in accordance with a first exemplary embodiment of the present invention. 
         FIG. 8  is a line plot of the mode input and the run input signals received by the SMPS health monitor of  FIG. 7 , in accordance with the first exemplary embodiment of the present invention. 
         FIG. 9  is a line plot of the mode-inject control and the frequency-gate control signals received by the SMPS health monitor of  FIG. 7 , in accordance with the first exemplary embodiment of the present invention. 
         FIG. 10  shows two significant types of damped ringing responses, fixed frequency (A) and variable frequency (B), in accordance with the first exemplary embodiment of the present invention. 
         FIG. 11  is a schematic diagram of the ring frequency detector for the SMPS health monitor shown in  FIG. 7 , in accordance with the first exemplary embodiment of the present invention. 
         FIG. 12  is a line plot of the analog and the digitized result of the output of the ring frequency detector for a fixed-frequency damped ringing response for a “no degradation” and an “intermediate degradation” condition. 
         FIG. 13  is a line plot of the analog and the digitized result of the output of the ring frequency detector for a variable-frequency damped ringing response for a “no degradation” and an “intermediate degradation” condition. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     An abrupt current change in an SMPS, such as that caused by a sudden change in the effective load resistance, will result in a damped ringing response on the output voltage.  FIG. 3  is a line plot showing a ringing response, such as that in response to an abrupt change in current, on a 5.0 V output voltage of a SMPS. An SMPS has at least one direct voltage  41  as an output, which in this example is 5.0 V. At time 250 microseconds, an abrupt change in current occurs, which causes a damped ringing response. The damped ringing response begins with a ringing voltage change  43 . The damped ringing response is a series of sinusoidal waves  45  at a constant resonant frequency and decreasing amplitude. The damped ringing response is almost completely damped at time around 500 microseconds in this exemplary embodiment, after which the SMPS again outputs a direct voltage of 5.0 V. 
     The SMPS output shown in  FIG. 3  can be modeled by the following expression: (EQ. 1): V O =V DC +A R {exp(−t/τ)}{cos(ωt+φ)} where V DC  is the direct voltage  45  output of the SMPS; A R  is the peak amplitude of the dampened ringing response; t is time; τ is the dampening time constant; (EQ. 2): ω=2πf R  where f R  is the resonant frequency of the dampened ringing response; and φ is the phase shift of the resonant frequency. The terms A R , τ and ω are complex expressions dependent primarily upon the exact topology of the SMPS, especially the feedback loop; the current mode of the SMPS (continuous current flow or discontinuous current flow); and the type of the abrupt current change (impulse or step). For continuous current mode and an impulse type of current change, the terms become the following: 
                             A   k     =       -     Z   out       ⁢   Δ   ⁢           ⁢   I   ⁢           ⁢     A     A   +   1       ⁢     (     1       ω   0     ⁢       1   -     1     4   ⁢     Q   2                 )         ,           (     EQ   .           ⁢   3     )                 τ   =     (       2   ⁢   Q       ω   0       )       ,           (       EQ     .           ⁢   4     )                 ω   =       ω   0     ⁢       1   -     1     4   ⁢     Q   2                 ,           (       EQ     .           ⁢   5     )                 ω   0     =         A   +   1       ⁢     (     1     LC       )     ⁢           ⁢   and             (       EQ     .           ⁢   6     )               Q   =         A   +   1       ⁢       (       1   R     ⁢       C   L         )     .               (       EQ     .           ⁢   7     )                 
The derivation of these expressions and terms are not pertinent to the present invention: what is pertinent is they show the amplitude, the duration of the ringing and the frequency of the ringing response are related to and dependent upon the gain (A), resistance (R), capacitance (C) and inductance (L) of the feedback loop of the SMPS.
 
     As indicated by these equations and expressions, there are three variables that change in response to an abrupt stimulus such as an abrupt change in load current: amplitude, dampening time and ring frequency. Of these three, the dampening time and the ring frequency are particularly amenable to prognostication methods. 
       FIG. 4  is a line plot of the loop gain  51  (A) in dB and the phase  53  (B) in degrees of an exemplary SMPS versus frequency. The crossover frequency  55  is the frequency (f C ) at which the loop gain  51  is 0 dB. For the SMPS to be stable, the phase margin  57  (180 degrees minus the absolute value of the phase) must be positive and greater than some design margin (for instance, 45 degrees). The SMPS represented by the plots in  FIG. 4  is stable because the phase margin  57  is greater than 45 degrees. The SMPS has a resonant frequency, f R,  at point  59  of the phase plot. Resonant frequency is the frequency at which the phase is minus 180 degrees. The SMPS does not oscillate because the gain margin  61  is less than 0 dB at the resonant frequency of the SMPS (gain margin  61  is herein defined as the loop gain  51  at the frequency in which the phase  53  is minus 180 degrees). An abrupt change, such as that induced by an abrupt change in the load current, introduces disruptions that cause the SMPS to begin to oscillate: a ringing response. The SMPS exhibits a damped ringing in response to an abrupt current change because the gain is less than 1 (negative value in dB). 
       FIG. 5  is a schematic diagram of a first exemplary embodiment of a feedback circuit for a SMPS. The feedback circuit  117  includes input terminals  115 A and  115 B connected to positive and negative nodes of an output voltage of a SMPS. Output terminals  119 A and  119 B are connected to a PWM, such as the PWM shown in  FIG. 1 . Opto-isolator  172 , an amplifier, is connected to a load resistor  174 , which is connected to the output of a feedback amplifier  180 , which may be an operational amplifier. A first input of the feedback amplifier  180  is connected to a voltage divider comprising two resistors  182  and  184  and a second input of the feedback amplifier  180  is connected to an input network including resistors  186 ,  188 ,  190 , a first capacitor  192 , a second capacitor  194 , and a biasing voltage  196 . The values of the resistors  186 ,  188 ,  190  and the biasing voltage  196  determine the overall gain of the feedback amplifier  180 . The gain of the feedback amplifier  180  and the gain of the opto-isolator  172  determines the gain of the feedback circuit  117 . The capacitors  192 ,  194  values are chosen to ensure the gain  51  and the phase  53  (both shown in  FIG. 4 ) result in a stable operation. A degradation of gain of the feedback circuit  117  is almost always because of a degradation of gain of the opto-isolator  172  and not because of degradation in the gain of the feedback amplifier  180 . 
       FIG. 6  is an exemplary current injection device  134 , in accordance with a first exemplary embodiment of the present invention. The current injection device  134  includes an injection inverter  136 , an injection switch  138  having a P-channel power MOSFET and a injection load resistor  140 . The current injection device  134  has a first terminal  142  that is connected to a voltage output bus  104  at first connection point  106 . A second terminal  144  is connected to terminal  115 A or terminal  115 B (shown in  FIG. 5 ). When an input  146 , connected to the injection inverter  136 , is positive, the injection inverter  136  turns on the injection switch  138  to connect the injection load resistor  140  to the voltage output bus  104  at first connection point  106 . The result of the positive input  146  is an abrupt change in the load current of the SMPS (such as the prior art SMPS shown in  FIG. 1 ), which results in a damped ringing such as that shown in  FIG. 3 . The current injection device  134  is able to inject an abrupt current change of known duration and it injects a current change of known amplitude, which is the output voltage at the voltage output bus  104  divided by the value of the injection load resistor  140 . 
       FIG. 7  is a block diagram of a SMPS health monitor  120  connected to the output voltage bus  104  of a SMPS  101 , in accordance with a first exemplary embodiment of the present invention. The SMPS  101  has a voltage output terminal  102 , which can be either the positive or the negative terminals  115 A,  115 B shown in  FIG. 5  (in the first exemplary embodiment, a positive output terminal is shown). The voltage output terminal  102  connects to the output voltage bus  104  to which are attached electronic assemblies at the first connection point  106 . The SMPS health monitor  120 , which has a mode input  122 A, a run input  122 B, a filter selection input  124  and a digital output  118 . The SMPS health monitor  120  also includes digital logic  126  having a prognostic control  128 , a counter  130 , and a digital filter  132 . SMPS Health Monitor  120  also includes the current injection device  134  and an analog input frequency detector  110 . The prognostic control  128  operates in at least one of two modes, monitor or self-test, as may be determined by the mode input  122 A. 
     Still referring to  FIG. 7 , when the run input signal  122 B is received and the prognostic control  128  is operating in monitor mode, the prognostic control  128  sends an inject control  112  to the current injection device  134 , which causes an abrupt current change at the first connection point  106 . Then the prognostic control  128  sends a frequency-gate control  114  to the frequency detector  110 , which detects the analog negative swings of the damping ringing response. The detected analog negative swings are amplified by the frequency detector  110  and sent as digital pulses to the counter  130 , which counts the pulses. The count of the number of pulses caused by the damping ringing response is sent to the digital filter  132 , which filters the digital count to produce a prognostic output  118 . The prognostic output  118  indicates the level of degradation of the feedback loop in the SMPS  101 , with the degradation most likely being due to degradation in CTR of the opto-isolator  172  in the feedback loop. A clock input  125  is used by digital logic  126 . 
       FIG. 8  is a line plot of the mode input  122 A and the run input  122 B signals received by the SMPS health monitor  120  of  FIG. 7 , in accordance with the first exemplary embodiment of the present invention. The mode input  122 A has two values: when mode is positive  150 , the SMPS health monitor  120  is in monitor mode and will inject an abrupt current change to the SMPS  101  whenever the run input  122 B is a positive pulse  154 . When mode is initially pulled low  152 , the SMPS health monitor  120  is placed in self-test mode and the prognostic control  128  sends a reset signal to the counter  130 . Whenever run input  122 B is a positive pulse  156  and the mode is self-test, the counter  130  is incremented by one. The self-test mode enables verification of correct operation of the counter  130  and the digital filter  132 . 
       FIG. 9  is a line plot of the mode-inject control and the frequency-gate control signals received by the SMPS health monitor of  FIG. 7 , in accordance with the first exemplary embodiment of the present invention. The positive pulse  154  of run input  122 B in  FIG. 8(B)  causes the prognostic control  128  to generate the inject control  112 , which is the first control pulse  160 . The prognostic control  128  also generates the frequency-gate control  114  as shown by second control pulse  162 . 
       FIG. 10  shows two significant types of damped ringing responses, fixed frequency (A) and variable frequency (B), in accordance with the first exemplary embodiment of the present invention. When an opto-isolator  172  degrades, an abrupt current change causes one of two significantly different types of fault-to-failure progression (FFP) signatures. The first type of FFP signature is characterized by a significant change in the dampening duration of the response as seen by the difference in a longer response  180  compared to a shorter response  182 . The difference is predicted by equations EQ. 1, EQ. 4 and EQ. 7. There is a significant difference in the number of detectable cycles between the longer response  180  and the shorter response  182 . A reduction in amplitude, as predicted by equations EQ.1 and EQ. 3, is also seen, and such amplitude reduction also contributes to the reduction in the number of detectable cycles of ringing. 
     The second type of FFP signature is characterized by a significant reduction in the frequency of the ringing as seen by the difference in a quicker response  184  compared to a slower response  186 . The change in frequency is predicted by equations EQ. 1, EQ. 5 and EQ. 6. As seen comparing the quicker response  184  and the slower response  186 , there is a significant difference in the number of detectable cycles of ringing. Whether there is or is not also a change in dampening time or ringing amplitude is not significant, because any such change also contributes to the reduction in the number of detectable cycles of ringing. This reduction in the number of detectable cycles of ringing is directly related to a reduced gain in the feedback loop, and the gain reduction is likely due to a reduced CTR of the opto-isolator  172 , and the reduced CTR is due to degradation in the opto-isolator  172 . The exploitation of the reduction in the number of detectable cycles of ringing frequency due to a reduction in amplitude and/or frequency and/or dampening time, all which are the result of reduced gain, as the opto-isolator  172  in the feedback loop degrades is an important element of the present invention. 
       FIG. 11  is a schematic diagram of the frequency detector  110  for the SMPS health monitor  120  shown in  FIG. 7 , in accordance with the first exemplary embodiment of the present invention. The frequency detector  110  includes a differential input amplifier  188 , a detector resistor  190 , a detector capacitor  192  and an AND logic gate  196 . The frequency detector  110  is attached to the first and second terminals  142 ,  144  of the current injection device  134  (shown in  FIG. 6 ). The detector resistor  190  and the detector capacitor  192  filters out the damped ringing frequency on the direct voltage at the first connection point  106  of voltage output bus  104  to provide a direct voltage to the positive input of the differential input amplifier  188 . This filtering and the direct connect of the negative input of the differential input amplifier  188  to the first connection point  106  results in the differential input amplifier  188  outputting positive pulses corresponding to the negative swings of the sinusoidal waves in the damped frequency response. The output  194  of the differential input amplifier  188  is ANDed by the AND logic gate  196  and the frequency gate control  114 . The output  194  of the differential input amplifier  188  to the ringing frequency responses  180 ,  182  shown in  FIG. 10A , is shown in  FIG. 12A  and the output  198  of the AND logic gate  196  is shown in  FIG. 12B . The output  198  is sent to the counter  130  (shown in  FIG. 7 ), which results in a count of 8 and a count of 4 being input to the digital filter  132  (shown in  FIG. 7 ). Similarly, the output  194  of the differential input amplifier  188  to the ringing frequency responses (shown in  FIG. 10B ) is shown in  FIG. 13A  and the corresponding output  198  of the AND logic gate  196  is shown in  FIG. 13B . The output  198  of the AND logic gate  196  is sent to the counter  130  (shown in  FIG. 7 ), which results in a count of 8 and a count of 5 being input to the digital filter  132  (shown in  FIG. 7 ). The digital filter  132  would produce a prognostic health signal corresponding to no degradation when the count is maximum, maximum degradation when the count is 0 or 1, and intermediate levels of degradation for counts between the maximum and a count of one. 
     The digital logic  126  (shown in  FIG. 7 ) is easily realized by using a hardware description language (HDL), such as Verilog, to program a field programmable gate array (FPGA). The implementation of the digital logic  126  is not described herein, although one having ordinary skill in the art would be able to implement the digital logic  126  herein described without undue experimentation. The digital logic  126  may include the methods, as previously described, to control and sequence the abrupt current change, the gating of the frequency detector to produce pulses corresponding to frequency cycles in the damped frequency response, the counting of those pulses and the outputting of prognostic health signal levels to indicate no degradation, maximum degradation and intermediate levels of degradation. These output signal levels provide a prognostic progression from no degradation to maximum degradation; the prognostic progression plus evaluation of the times it takes to progress from one level to the next provide the basis for producing accurate RUL estimates. 
     The digital logic  126  can then be implemented by synthesizing a Verilog program and loading the synthesized program into, for example, a Field Programmable Gate Array or by using the synthesized output to create a digital schematic that can be implemented as an integrated circuit or as a collection of discrete digital gates. Referring to  FIG. 7 , an exemplary behavioral level of a program is the following:
         1. When the mode input  122 A is at a positive edge, send a digital reset to the counter  130  and pull the controls  112 ,  114  inactive low.   2. When the mode input  122 A is at a negative edge, send a digital reset to the counter  130 .   3. When the mode input  122 A is active high and a positive pulse is received at the run input  122 B, send a positive pulse through inject control  112  to the current injection  134  to cause the injection of an abrupt change in SMPS  101  load current. Also send a reset pulse to the counter  130 , also send a positive frequency-gate control  114  to the frequency detector  110 , with the length of the frequency-gate control  114  being for a predetermined number of digital clock cycles, with the number corresponding to the expected maximum dampening time of the damped ringing response.   4. When the mode input  122 A is inactive low and a positive pulse is received at the run input  122 B, send a test increment count signal to the counter  130  to create an increasing test count to the digital filter  132 .   5. For each digital value of the input bits  124  to the prognostic control  128 , enable a predetermined number of clocks cycles to control the width of the frequency-gate control  114 .   6. For each digital value of the input bits  124  to the digital filter  132 , enable a corresponding set of combinational logic gates to transform a count from the counter  130  to a predetermined prognostic health signal.   7. Define a set of combinational logic gates for each supported SMPS  101  or group of SMPS  101 .       

     It should be emphasized that the above-described embodiments of the present invention, particularly, any “preferred” embodiments, are merely possible examples of implementations, merely set forth for a clear understanding of the principles of the invention. Many variations and modifications may be made to the above-described embodiments of the invention without departing substantially from the spirit and principles of the invention. All such modifications and variations are intended to be included herein within the scope of this disclosure and the present invention and protected by the following claims.