Abstract:
In one embodiment, a dynamic random access memory (DRAM) is provided that includes: a plurality of rows of memory cells, each of the memory cell rows being arranged into columns, wherein each of the memory cell rows is crossed by a row of four word lines, and wherein each of the columns is crossed by a bit line; a plurality of sense amplifiers corresponding to the bit lines such that a single sense amplifier corresponds to every four bit lines; and a plurality of 4:1 multiplexers corresponding to the plurality of sense amplifiers, each 4:1 multiplexer coupling its corresponding sense amplifier to its corresponding four bit lines.

Description:
RELATED APPLICATION 
     This application is a continuation-in-part of U.S. application Ser. No. 11/370,738, filed Mar. 7, 2006, now U.S. Pat. No. 7,366,046 which in turn claims the benefit of U.S. Provisional Application No. 60/708,729, filed Aug. 16, 2005, the contents of both applications being incorporated by reference in their entirety. 
    
    
     BACKGROUND OF THE INVENTION 
     The present invention relates to memories, and more particularly to a dynamic random access memory (DRAM) architectures with increased densities. 
     Because a DRAM memory cell includes just a single access transistor and a storage capacitor, DRAM offers dramatic density advantages over static random access memory (SRAM), which typically requires a six transistor (6-T) memory cell. In contrast to SRAM, a DRAM cell will only change the voltage on its corresponding bit line slightly during read operations. Having been coupled to the bit line, the storage capacitor in a DRAM cell must be then be restored after a read operation. Thus, DRAM sense amplifiers require a regenerative latching ability to drive the bit line “full rail” after a read operation. If the sense amplifier determines that the storage capacitor was charged to VDD, the bit line is then driven to VDD to restore the charge on the storage capacitor. On the other hand, if the sense amplifier determines that the storage capacitor was not charged, the bit line is grounded to discharge the storage capacitor. Moreover, the charge on DRAM storage capacitors continually leaks away, requiring constant refreshing. SRAM cells require no such refreshing. In addition, because the 6-T SRAM cell can drive its value onto a bit line during read operations, SRAM is generally substantially faster than DRAM. 
     As a result of the density vs. speed advantages of SRAM and DRAM, SRAM is faster but more expensive and thus reserved for more time-critical operations such as a microprocessor&#39;s cache. To reduce costs, the remaining RAM for a microprocessor is then typically implemented as DRAM. However, because DRAM operation speed is constantly being improved, the use of embedded DRAM in integrated circuits is becoming more popular in high-performance applications that traditionally demanded embedded SRAM. Nevertheless, the choice between DRAM and SRAM is often guided by the density vs. speed tradeoffs discussed above. Accordingly, designers strive to increase DRAM density. 
     The challenges to increasing DRAM density may be better appreciated through discussion of a conventional DRAM  100  illustrated in  FIG. 1 . A word line WL 0  controls the gate of an access NMOS transistor M 0  whose drain couples to a bit line Bx. The source of access transistor M 0  couples to a storage capacitor C 0 . Thus, access transistor M 0  and storage capacitor C 0  form a DRAM memory cell that is accessed by raising the voltage on word line WL 0 . In response to this voltage rise, any charge on capacitor C 0  couples to bit line Bx. A sense amplifier reads the contents of the accessed memory cell by detecting a resulting voltage change on bit line Bx. 
     The sense amplifier detects the voltage change by comparing the voltage on bit line Bx to a neighboring bit line such as a bit line B. Before this comparison is made, bit lines Bx and B are pre-charged to a voltage VDD/2 by pre-charge circuitry. If the comparison indicates that bit line Bx is higher in voltage than bit line B, downstream decoding logic (not illustrated) will assume that the storage capacitor C 0  had previously been charged to VDD. If the comparison indicates that bit line B is higher in voltage than bit line Bx, the decoding logic will assume that storage capacitor C 0  had previously been discharged. In this fashion, a decision is made as to the binary contents of the memory cell. Having read the contents of the memory cell, the sense amplifier will restore the memory cell using a regenerative latch. An analogous access may be made to a memory cell comprised of access transistor M 1  and a storage capacitor C 1  by raising a word line WL 1 , and so on. 
     Each bit line has an inherent capacitance that is typically an order of magnitude greater than the capacitance of the storage capacitors. This difference in capacitance is exacerbated as the number of memory cells accessible by a sense amplifier is increased. For example, should DRAM  100  be first implemented with an integer number “N” of word lines, the length of the bit lines will have to double if N is increased to 2*N (assuming the same semiconductor process dimensions in both cases). The bit line capacitance will thus double as well, thereby decreasing the voltage change when a memory cell is accessed. As a result, the maximum number of memory cell rows per sense amplifier in a conventional trench-capacitor DRAM is limited to, for example, 512 rows per sense amplifier. The maximum number of memory cell rows per sense amplifier is much lower if storage transistors are used to form the memory cells such as a maximum of 8 to 16 rows. 
     One of the limiting factors in sensing ever-smaller voltage changes as DRAM density is increased is the non-ideal characteristic of sense amplifiers. A conventional sense amplifier  200  is illustrated in  FIG. 2 . This sense amplifier includes a differential amplifier  205  that compares a voltage P derived from bit line B ( FIG. 1 ) at its positive input to a voltage N derived from bit line Bx ( FIG. 1 ) at its negative input. If voltage P is higher than the voltage N, the voltage difference is amplified onto differential outputs P 0  and N 0  by the differential amplifier&#39;s gain. A regenerative latch  210  would then drive output P full rail to VDD and ground output N. Conversely, if voltage N is higher than voltage P, the regenerative latch drives output N full rail to VDD and grounds output P. 
     Should both differential inputs, however, be at the same voltage (such as the pre-charge voltage VDD/2), the regenerative latch operation just described may not take place correctly. Instead, because of offset imperfections in the differential amplifier, P 0  (for example) output may be driven higher than N 0  despite the equal voltages at the inputs. In turn, this offset limits the sensitivity of the sense amplifier operation. For example, suppose bit line B should be higher in voltage than bit line Bx during a read operation. If the storage capacitance is too small with regard to the bit line capacitance, the offset within the differential amplifier may drive the regenerative latch to pull output N full rail, leading to an erroneous reading. 
     It may be seen that other issues affect sense amplifier decisions. For example, suppose both bit lines are discharged to VSS (ground potential) prior to sensing. Such a scenario favors the sensing of charged memory cells as opposed to discharged memory cells. Conversely, if both bit lines are charged to VDD, the sensing of discharged memory cells is favored over charged memory cells. The favoring of one bit decision (declaring a binary one or a binary zero as the memory cell contents) over another may be exacerbated as density increases. Accordingly, there is a need in the art for improved DRAM architectures that address such common-mode effects on bit decisions. 
     As discussed above, the storage capacitance vs. bit line capacitance is a limiting factor for DRAM density. By increasing the storage capacitance, a sense amplifier can better decide what binary contents are being stored. However, the increased storage capacitance generally leads to increased memory cell size, thereby diminishing density. Accordingly, there is a need in the art for improved DRAM memory cell architectures that maximize achievable density. 
     Another factor affecting density is power consumption. Because DRAMs perform destructive reads, the accessed cell contents must be restored after read operations. In general, all memory cells in a row coupled to a word line will be accessed as that word line is asserted. To limit the number of memory cells that must be restored upon any given word line assertion, it is known to use even and odd word lines such that each row of memory cells associates with two word lines. However, such a memory will still access half the memory cells in a row upon the assertion of a word line. As memory densities increase, such an architecture will still suffer from appreciable power losses during read (and refresh) operations. Accordingly, there is a need in the art for improved DRAM architectures that limit the power consumption during read and refresh operations. 
     SUMMARY 
     This section summarizes some features of the invention. Other features are described in the subsequent sections. 
     In accordance with a first aspect of the invention, a dynamic random access memory (DRAM) is provided that includes: a plurality of rows of memory cells, each of the memory cell rows being arranged into columns, wherein each of the memory cell rows is crossed by a row of four word lines, and wherein each of the columns is crossed by a bit line; a plurality of sense amplifiers corresponding to the bit lines such that a single sense amplifier corresponds to every four bit lines; and a plurality of 4:1 multiplexers corresponding to the plurality of sense amplifiers, each 4:1 multiplexer coupling its corresponding sense amplifier to its corresponding four bit lines. 
     In accordance 
     The invention is not limited to the features and advantages described above. Other features are described below. The invention is defined by the appended claims. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  illustrates a conventional DRAM; 
         FIG. 2  illustrates a conventional sense amplifier; 
         FIG. 3  illustrates a sense amplifier incorporating a self-bias generation circuit in accordance with an embodiment of the invention; 
         FIG. 4  illustrates a self-bias generation circuit in accordance with an embodiment of the invention; 
         FIG. 5  illustrates a DRAM having multiple-word-lines-per-memory-row in accordance with an embodiment of the invention; 
         FIG. 6  illustrates a multiplexer and pre-charging circuit in accordance with an embodiment of the invention; 
         FIG. 7  illustrates a DRAM memory cell layout in accordance with an embodiment of the invention; 
         FIG. 8  is a cross-sectional view of DRAM storage capacitors formed in the metal layers adjacent a substrate in accordance with an embodiment of the invention; 
         FIG. 9  is a plan view of a metal layer for the DRAM storage capacitors of  FIG. 8  in accordance with an embodiment of the invention; 
         FIG. 10  illustrates the via arrangement for a metal layer of  FIG. 8  in accordance with an embodiment of the invention; 
         FIG. 11  illustrates another via arrangement for a metal layer of  FIG. 8  in accordance with an embodiment of the invention; and 
         FIG. 12  is a cross-sectional view of DRAM storage capacitors having the via arrangement of  FIG. 11 . in accordance with an embodiment of the invention. 
         FIG. 13  illustrates a DRAM word line arrangement that permits a 4:1 local multiplexing of the word lines at the sense amplifiers in accordance with an embodiment of the invention. 
     
    
    
     DETAILED DESCRIPTION 
     Reference will now be made in detail to one or more embodiments of the invention. While the invention will be described with respect to these embodiments, it should be understood that the invention is not limited to any particular embodiment. On the contrary, the invention includes alternatives, modifications, and equivalents as may come within the spirit and scope of the appended claims. Furthermore, in the following description, numerous specific details are set forth to provide a thorough understanding of the invention. The invention may be practiced without some or all of these specific details. In other instances, well-known structures and principles of operation have not been described in detail to avoid obscuring the invention. 
     Turning now to  FIG. 3 , an improved sense amplifier  300  includes a self-bias generation circuit  305 . The self-bias generation circuit complements the operation of a pre-charge circuit  310  that operates to charge a bit line B and a bit line Bx to VDD/2 prior to a read operation. Should the pre-charge circuit operate without the self-bias generation circuit, a differential amplifier  315  will, due to its offset, provide some voltage difference between a positive output P 0  and a negative output N 0  even though bit lines B and Bx are at the same voltage. To cancel this voltage difference, the voltages of bit lines B and Bx may be adjusted in a complementary fashion from VDD/2 so that the voltages of P 0  and N 0  are equal. In this manner, the differential amplifier may then immediately respond to any additional changes in the voltages of bit lines B and Bx due to a read operation by amplifying the additional change according to its internal gain. This amplified change is then applied across outputs P 0  and N 0  so that a regenerative latch  320  may drive its outputs P and N accordingly. Outputs P and N then couple back (coupling not illustrated) to the bit lines so that the appropriate voltage is restored on the storage capacitor that was just accessed. 
     An exemplary self-bias generation circuit  400  is illustrated in  FIG. 4 . Operation of circuit  400  may be isolated from the pre-charge operation. In other words, the pre-charge circuit of  FIG. 3  charges the bit lines B and Bx to VDD/2 and then lets these lines float while the self-bias generation circuit operates to cancel the offset in the differential amplifier. Similarly, the self-bias generation circuit operation may be isolated from the raising of the word line voltage discussed with regard to FIG.  1 —without such isolation, the voltage difference from the storage capacitor being accessed would be treated as an offset by the self-bias generation circuit and thereby erased. As seen in  FIG. 4 , transmission gates  405  and  410  are controlled by an active low signal nfbx. When signal nfbx is asserted (pulled low), transmission gate  405  becomes conductive such that the voltage of input P equals the voltage of output N 0 . Similarly, transmission gate  410  becomes conductive such that the voltage of input N equals the voltage of output P 0 . Referring again to  FIG. 3 , it may be seen that the resulting negative feedback through the differential amplifier dramatically reduces the effects of the voltage offset. For example, if the voltage offset is represented by ΔV, the relationship between the input and output voltages is given by:
 
 G ( V   P   −V   N )+Δ V=V   P0   −V   NO  
 
where G is the gain of the differential amplifier, V P  and V N  are the voltages at the input nodes P and N, respectively, and V PO  and V NO  are the voltages at the output nodes P 0  and N 0 , respectively. Should V PO  equal V N  and V NO  equal V P  such as when the signal nfbx is asserted, the preceding equation leads to:
 
Δ V /( G+ 1)=−( V   P   −V   N )
 
As a result, the offset is reduced by the factor (G+1). Because the open loop gain G is large for a differential amplifier, the effect of the offset voltage is virtually eliminated. It will be appreciated that other topologies may be used for the self-bias generation circuit to eliminate the offset effect in this fashion.
 
     Although the sense amplifier discussed with regard to  FIG. 3  advantageously increases the sensitivity of its differential amplifier, the resulting biases may affect the margins regarding whether an accessed memory cell was storing a “1” or a “0.” For example, as discussed with regard to the conventional sense amplifier of  FIGS. 1 and 2 , the bit lines are pre-charged to VDD/2. Such a bias neither favors a logical 1 nor a logical zero decision and is thus theoretically ideal. However, the self-bias generation circuit described herein will pull the bit. lines away from this ideal bias point such that a logical 0 may be favored over a logical 1 decision (or vice versa). In general, the ideal bias point for a given differential amplifier will depend upon the relative strengths of its PMOS and NMOS transistors. In turn, these relative strengths are affected by the particular semiconductor process corner used to manufacture the transistors. A customer of a semiconductor foundry can never guarantee, a priori, what particular process corner will be used to manufacture a given batch of wafers. Thus, the relative margin between logical 0 and logical 1 decisions for a sense amplifier that includes a self-bias generation circuit may also be unpredictable. 
     As a result, although a sense amplifier with self-bias generation advantageously is more sensitive to the voltage differences produced by accessing a memory cell, the self-bias generation may push the sense amplifier into having less than ideal margins between logical 0 and logical 1 decisions. Referring again to  FIG. 3 , bit line B may include a trim circuit such as a trim capacitor  325  charged by an adaptive signal (trimx) such that a less-than-ideal margin may be optimized. Similarly, bit line Bx may include a trim capacitor  330  charged by an adaptive signal (trimy) to also optimize the margin. Upon manufacture of a DRAM having sense amplifiers with self-bias generation, a user may test the bit error rate for storing the binary states (logical 0 vs. logical 1). Should the bit error rate for storing, for example, a logical 0 in each memory cell be higher than when a logical 1 is stored, the adaptive signals trimx and trimy may be adjusted such that the bit error rate is minimized for both decisions. During idle states, the adaptive trim signals are not asserted. The adaptive trim signals may range through a binary-weighted range of values. Upon manufacture, the appropriate trim signal values to be applied after self-bias generation may be “burned” into the circuit using, for example, fuses or non-volatile memory. It will be appreciated that a trim circuit may be used in other memories such as SRAM or flash. For example, the trim circuit may comprise an adjustable current source should a memory use current sensing to make bit decisions. Regardless of the type of bias adjustment provided by a trim circuit coupled to a bit line (such as charge from a capacitor or current from a current source), the trim circuit may be adjusted by a controller depending upon the observed bit error rate to enhance performance. Moreover, a trim circuit may be used during debugging to test memory performance in response to trim-circuit-provided bit line bias adjustments. 
     The sense amplifier discussed with regard to  FIG. 3  advantageously increases the sensitivity to the voltage differences produced by accesses to storage capacitors in a DRAM, thereby increasing the achievable storage capacity for a given die area dedicated to the DRAM. However, the additional self-bias generation circuit increases the die area needed to implement the sense amplifier. Referring back to  FIG. 1 , it may be seen that a DRAM sense amplifier must typically fit within the pitch between neighboring bit lines. Such a pitch continues to decrease as modern semiconductor processes go into the deep sub-micron regime. Thus, it may be difficult to implement a sense amplifier with self-bias generation in such narrow pitches. 
     Turning now to  FIG. 5 , a DRAM having four word lines per memory cell row is illustrated that provides additional pitch space for a sense amplifier  510 . For illustration clarity, only three memory cell rows are shown, ranging from a zeroth row to a second row. Similarly, only four memory cells are shown per row, such as cell  0 - 0  in the zeroth row and a cell  2 - 3  in the second row. The four memory cells in each row are arranged in columns, each column having a bit line. Thus, a zeroth column has a bit line B 0  and so on such that a third column has a bit line B 3 . A multiplexer  505  selects between pairs of neighboring bit lines for processing by a sense amplifier with self-bias generation  510 . Each row of memory cells has four word lines. For illustration clarity, only word lines W 0  through W 3  are illustrated for row  1 . Each word line only couples to one memory cell in each group of four memory cells. For example, word line W 0  couples to the zeroth cell in row  1 . Word line W 1  couples to the first cell in row  1 . Word line W 2  couples to the second cell in row  1 . Finally, word line W 3  couples to the third cell in row  1 . 
     The multiplexer is controlled such that if the zeroth cell in row  1  is to be accessed, bit lines B 0  and B 1  are selected for processing. Similarly, these same lines are selected if the first cell in row  1  is to be accessed. However, if either the second or third cells in row  1  are to be accessed, bit lines B 2  and B 3  are selected for processing. Because only one cell out of each group of four in each row is accessed at any given time, multiplexer  505  acts as a 4:1 multiplexer. However, in other embodiments, the multiplexing may differ such as an 8:1 multiplexer. It will be appreciated, however, that an 8:1 multiplexing would require 8 word lines per row. Additional density may be achieved by duplicating the multiplexer and DRAM structure on the other side of the sense amplifier as indicated by the arrow in  FIG. 5 . 
     Because four word lines cross each row of cells in DRAM  500 , the word lines may twist across metal layers to minimize coupling between adjacent word lines. For example, an example word line arrangement is shown in  FIG. 13 . For illustration clarity, only a single row of word lines is shown traversing a core array of memory cells. In this single row, a word line  0  and a word line  1  couple to a wordline right driver  1305  using a first semiconductor process metal layer such as a metal layer  3 . In contrast, a word line  2  and word line  3  couple to a wordline left driver  1310  using a second semiconductor process metal layer such as metal layer  2 . Because such multiple word lines may disadvantageously suffer from capacitive coupling such that the assertion of one of the word lines would couple to other word lines in the same row, the word lines are laced through to adjacent metal layers through a plurality of switching points. In this fashion, two of the word lines such as word line  1  and word line  3  may remain within their respective metal layers. However, the remaining word lines such as word lines  0  and word line  2  switch their respective metal layers at the switch points through vias. For example, at a first switch point, word line  0  switches from metal layer  3  to metal layer  2  whereas word line  2  switches from metal layer  2  to metal layer  3 . It may be appreciated there should be an even number of switch points such that a mirror image of the 1 st  switch point exists at a second switch point where word line  0  switches back to its “home” metal layer  3  and word line  2  switches to its metal layer  2 . In this fashion, capacitive coupling between the word lines in a row is minimized. It will be appreciated that the 4:1 multiplexing that is thereby achieved offers significant power savings over the customary even/odd multiplexing. For example, suppose there are 1024 memory cells per row. In an even/odd word line arrangement, 512 memory cells would be accessed upon the assertion of any given word line. However, by implementing a 4:1 local multiplexing at the sense amplifier as discussed herein, only 256 memory cells would be accessed in a read operation, thereby reducing the amount of power wasted during the DRAM destructive read process in half. 
     To achieve better layout, the word line drivers for the word lines may be separated on either side of the array as shown in  FIG. 13 . To prevent coupling to dormant word lines, the word lines corresponding to the inactive word line driver side may be strapped to ground through strapping transistors. For example, suppose it is word line right driver  1305  that is active (thereby asserting either of word line  0  or word line  1 ). The word lines in the remaining pair may be strapped to ground through strapping transistors  1320  and  1325 . 
     The multiplexer may also include pre-charging circuitry. An exemplary multiplexer and pre-charging circuit  600  is illustrated in  FIG. 6 . For illustration clarity, only one half of circuit  600  is illustrated, the remaining half being symmetric. Thus, just a first bit line b[0:1] and a second bit line bx[0:1] are shown. Each bit line is pre-charged when a selection signal SEL is not asserted (active high). Because signal SEL(0:1) is inverted at a first inverter  605 , it will turn on transistors  610  and  615  when this signal is not asserted. Thus each bit line will pre-charge to VDD/2, the theoretically ideal bias point discussed previously. To assure balance, a balance transistor  620  also conducts at this time to equalize the charge on each bit line. When signal SEL(0:1) is asserted, it will cause a transmission gate T 1  and T 2  to conduct because of the re-inversion at a second inverter  625 . The pre-charged bit lines will then couple to input nodes P and N discussed previously. 
     After processing the voltages on nodes P and N, the bit lines will be pulled full rail in a complementary fashion through the regenerative latching action of a sense amplifier (not illustrated). Suppose bit line b[0:1] is thus to be grounded. Referring again to  FIG. 1 , the corresponding access transistor may then leak, despite having its gate and source grounded. To reduce this leakage, the access transistor may be given a negative gate-to-source voltage (V gs ) by preventing the regenerative latch from fully grounding this bit line. To prevent this full ground, each bit line may couple through a weak transistor to VDD/2. For example, bit line b[0:1] couples through a weak transistor  630  whereas bit line bx[0:1] couples through a weak transistor  635 . Within each bit line pair, the gate of each weak transistor for a bit line is controlled by the voltage on remaining bit line. Because the bit lines are pulled full rail in a complementary fashion, should a bit line be pulled to ground by the regenerative latching action, its weak transistor will attempt to pull the bit line towards Vdd/2. In this fashion, through appropriate selection of the weak transistor dimensions, the access transistor for a “grounded” bit line may have a Vgs of, for example, approximately −200 to −300 millivolts, thereby greatly reducing leakage current from the corresponding column of inactive access transistors. 
     Referring again to  FIG. 5 , the use of four or more word lines per row provides a pitch of at least 4 bit lines in width for the sense amplifier. In this manner, sufficient die space to implement the self-bias generation in modern deep sub-micron semiconductor processes is enabled. 
     Advantageously, the DRAM memory and sense amplifier architecture discussed herein may be used to implement an embedded DRAM memory using conventional CMOS semiconductor processing techniques. Because such conventional techniques do not allow density-enhancing DRAM trench capacitor implementations, the storage capacitors in such embedded DRAM memories may be implemented using a thick gate oxide transistor. Despite the die area demands of using thick gate oxide transistors to implement the storage capacitors, density is enhanced because a sense amplifier with self-bias generation may service substantially more memory rows than a conventional embedded DRAM implementation. It will be appreciated, however, that the architectures discussed herein may also be used with other storage capacitors such as trench capacitors. 
     Because a transistor-implemented storage capacitor demands more die area than trench capacitors, the layout of the memory cells may be important to provide the most efficient use of available die area. Turning now to  FIG. 7 , an interlocking “L-shaped” memory cell architecture is illustrated for a four-word-line-per-memory-row DRAM  700 . The base of each L shape is formed by a thick field oxide transistor  705  acting as a storage capacitor. The access transistor forms the stem of each L shape. By coupling each L-shaped memory cell to an opposing L-shaped memory cell at a bit line contact via  710 , the coupled memory cells form a rectangular C shape. Although these coupled cells share a bit line, they are coupled to different word lines (not illustrated). The rectangular C shape is duplicated in a neighboring column in a mirror-image fashion such as seen in columns  715  and  720 . A pair of immediately-neighboring columns  725  and  730  are formed analogously but staggered with regard to columns  715  and  720  such that the L shapes in columns  720  and  725  interlock. In this fashion, die space is efficiently used for a 4-word-line-per-memory row architecture. It will be appreciated, however, that the interlocking L-shaped layout for memory  700  may be applied to DRAM architectures that do not incorporate multiple word lines per row. 
     Although the L-shaped layout for memory  700  advantageously increases density, the storage transistor in each memory cell still demands an appreciable die area. In addition, these storage transistors will continually conduct leakage current, a current loss that is exacerbated as transistor dimensions move into the deep sub-micron regime. Turning now to  FIG. 8 , a cross-sectional view of a DRAM storage capacitor  800  is illustrated using the conventional metal layers available in CMOS and other semiconductor processes. The number of metal layers available depends upon the particular semiconductor process being implemented is represented by an integer N. Thus, the metal layers range from a first metal layer M 1  separated from a substrate  805  by a first field oxide layer FOX 1  to a final metal layer MN separated from an underlying metal layer (not illustrated) by a field oxide layer FOXN. As seen also in the plan view of  FIG. 9 , each metal layer forms an inner node plate  805  surrounded by a mesh  810 , which acts as a common outer node to all the inner node plates. At least one via  820  joins inner node plates  820  to a source of an access transistor as described with regard to  FIG. 1 . 
     The common outer nodes in adjacent metal layers are joined through a plurality of vias  1000  as illustrated in  FIG. 10 . Advantageously, because a DRAM storage capacitor needs to conduct very little current, semiconductor fab design rules may be violated such that the common outer node forms a hollow cylinder about each stack of inner node plates. In other words, the vias are placed immediately adjacent each other and also the width of each via is adjusted so that it matches the width of the common outer node between adjacent inner node plates as seen in plan view in  FIG. 11  and cross-sectional view in  FIG. 12  (for illustration clarity, only two metal layers are shown in  FIG. 12 ). Because design rules may be violated to produce vias  1000  of such width, thereby resulting in voids internally to vias  1000 , such voids will have no real effect because of the very small currents that flow through a DRAM storage capacitor. Moreover, even if the capacitance provided by any one metal layer is not sufficient, the total capacitance provided by all the metal layers used to form storage capacitor  800  (for example, eight metal layers) is sufficient, even in high density (small separation between inner node plates) designs. 
     Increased density is not the only advantage of storage capacitor  800 . In addition, although field oxide is lower quality than the gate oxide used in a storage transistor, the field oxide layers are so much thicker than gate oxide thicknesses as to reduce leakage current as compared to a storage transistor implementation. 
     The above-described embodiments of the present invention are merely meant to be illustrative and not limiting. It will thus be obvious to those skilled in the art that various changes and modifications may be made without departing from this invention in its broader aspects. Therefore, the appended claims encompass all such changes and modifications as fall within the true spirit and scope of this invention.