Abstract:
A phase locked loop provides an output frequency that bears a fractional relationship to an input frequency and includes a controlled oscillator for generating the output frequency. The phase information is scaled in the amplitude domain to provide the fractional relationship.

Description:
FIELD OF THE INVENTION 
       [0001]    This invention relates to the field of phase locked loops (PLLs), and in particular to digital phase locked loops. 
       BACKGROUND OF THE INVENTION 
       [0002]    In PLL&#39;s it is quite common to use fractional relationships between the various frequencies, for example, for use with Forward Error Correction (FEC). The output frequency may be equal to M/N times the input frequency with M=255 and N=237. There are a few common methods to implement this fractional relationship, but those are in reality hampered by rather severe limitations in the final performance. 
         [0003]    In generic block diagrams there are two that typically will be used to generate an output frequency that is M/N times the input frequency. There are other implementations possible if M/N can be simplified by getting rid of common denominators of M and N, but that would of course be equivalent to rewriting the M/N fraction to a simpler fraction. 
         [0004]    The first block diagram shown in  FIG. 1  shows a pre-division by N, so that the 1/N of M/N is attained. The PLL has an M divider in the feedback, implying that the effective transfer of the loop only is M multiplication for frequency. Thus the output will run on M/N. 
         [0005]    The disadvantage of this approach is that the edges from input signal are not well applied. The phase detector does not receive every edge from the input signal, but only 1 out of every N signals (the divider blocks the other edges so that their precise edge information gets lost). This implies:
       a) The frequency into the phase detector is lower, and so must the maximum bandwidth of the PLL transfer be lower.   b) The reduced bandwidth is more difficult to implement. Typical filters will use a resistive element and a capacitive element, and a reduced bandwidth will typically increase the capacitor, up to a level where for instance integration is not feasible any more.   c) The reduced bandwidth will decrease the suppression of noise from charge pump and VCO.   d) In short: best overall jitter performance suffers from the N pre-division.       
 
         [0010]    In the second block diagram shown in  FIG. 2 , the division is done after the PLL. Now the PLL runs with more edges being applied to the PLL, so that in a general sense the quality of the PLL remains constant without extra design effort in the charge pump, filter, VCO and the like. However, the cost is now that the output of the VCO or CCO is now running on an N times higher frequency. The disadvantages are apparent:
       a) There is a fair chance the VCO frequencies are no longer feasible. For example, consider the case of an input frequency of 16.384 MHz, N equal to 237 and M equal to 255. The real output frequency is still low (about 17.628 MHz), but now the VCO is required to run 4177.92 MHz, which is not trivial in a standard CMOS process. Of course the numbers can easily become even more extreme.   b) Dividers that divide from the extremely high VCO frequency to lower frequencies become difficult to design.   c) The power will rise due to the higher frequency requirement. Depending on implementation this may reduce the quality. For instance in an integrated VCO the higher power consumption will influence other VCO&#39;s in the system.       
 
         [0014]    In short, both traditional systems have their problems, and it would be attractive to have an alternative technique that does not have same limitations. 
       SUMMARY OF THE INVENTION 
       [0015]    According to a first aspect of the invention there is provided a phase locked loop providing an output frequency that bears a fractional relationship to an input frequency, comprising a controlled oscillator for generating the output frequency; a feedback loop coupled to the output of the controlled oscillator and generating a feedback signal; a phase detector for comparing an input signal with the feedback signal, said phase detector providing phase information in the amplitude domain; and a scaling stage for scaling said phase information in the amplitude domain to produce said desired output frequency. 
         [0016]    In embodiments of the invention, a pulsed input signal is unsampled and all edges from the input signal are active in the phase detector (so no pre-division) and yet have limited frequencies out of the frequency generating element or controlled oscillator such a a VCO (Voltage Controlled oscillator). 
         [0017]    In one embodiment an input signal is applied to a MOD N counter, a feedback signal is applied to a MOD M counter, the outputs of said counters are applied to respective first and second scaling units, the first scaling unit applies an M/N scaling factor, and the outputs of said scaling units are applied to a subtractor, where N and M are integers. 
         [0018]    Embodiments of the invention perform scaling in the amplitude domain, rather than the time domain as in the prior art. This allows the PLL to have maximum bandwidth relative to the input frequency. Embodiments of the invention are able to input every edge of the input signal into the phase detector, unlike in prior art arrangements where only certain edges are input into the phase detector. This also avoids jitter problems that arise when performing such operations in the time domain. 
         [0019]    In another aspect the invention provides a phase locked providing an output frequency that bears a fractional relationship to an input frequency, comprising an input stage comprising a sampling unit, a counter and a decimator; and an output stage comprising a controlled oscillator for generating the output frequency; a feedback loop providing a feedback signal from the output of said controlled oscillator, and a scaling stage for scaling phase information to provide said fractional relationship, and wherein the input stage is implemented in hardware and the output stage is implemented in software. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0020]    The invention will now be described in more detail, by way of example only, with reference to the accompanying drawings, in which:— 
           [0021]      FIG. 1  is a block diagram of a PLL with N pre-division; 
           [0022]      FIG. 2  is a block diagram of a PLL with post-division; 
           [0023]      FIG. 3  is a block diagram of a digital PLL with sampled phase detector; 
           [0024]      FIG. 4  is a block diagram of a digital PLL with separate phase counters; 
           [0025]      FIG. 5  is a block diagram of a PLL with separate phase counters; 
           [0026]      FIG. 6  is a block diagram of a PLL with limited accuracy scaling on the input; 
           [0027]      FIG. 7  is a block diagram of a PLL with limited accuracy scaling on the input and feedback; 
           [0028]      FIG. 8  is a block diagram of a fractional PLL with an alternative block diagram; 
           [0029]      FIG. 9  is a block diagram of a fractional PLL with an alternative block diagram; 
           [0030]      FIG. 10  is a block diagram of a digital fractional PLL with scaling; 
           [0031]      FIG. 11  is a block diagram of a digital fractional PLL with partial software implementation; 
           [0032]      FIG. 12  is a block diagram of a digital fractional PLL with maximized software implementation; and 
           [0033]      FIG. 13  is a diagram illustrating a potential problem in an overflow situation with a sawtooth shaped phase signal. 
       
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       [0034]    In order to understand the invention, reference will be made to a subclass of PLL&#39;s is a digital PLL as shown in  FIG. 3 . In such a PLL, the incoming signal is first sampled in unit  12  before being applied to the phase detector  22 , which is in the form of an updown counter. In this case, the phase detector is a counter  22  that represents the phase difference up to an extended number of cycles. In the filter this phase difference may be processed, possibly with required decimation, and the PI operation (Proportional-Integral control). 
         [0035]    The size of the counter  22  forming the phase detector determines the maximum range of phase difference for the whole PLL if proper precautions are taken against overflows of that counter. The controlled oscillator  16  is a digitally controlled oscillator (DCO), which encompasses a whole class of known frequency generating elements. 
         [0036]    The feedback loop contains a divided-by-M  18  and the output of the DCO  16  is applied to the input of a divider-by-N  20 . 
         [0037]    The digital PLL can be slightly changed to perform the same function, but now in a slightly changed format as illustrated in the block diagram of  FIG. 4 . Here instead of detecting the phase difference with an up/down counter, the phase difference counter is split up in two up counters  22   a ,  22   b  and a subtraction unit  24  to produce the same phase difference. The output of the subtraction unit  24  is applied to the filter  14  in the same way as in  FIG. 3 . 
         [0038]    If for a moment the sampling is ignored, since this can be considered just to be a clock domain transition, the circuit of  FIG. 4  can be represented in equivalent analog terms as shown in  FIG. 5 . In this diagram the clock domains are not taken into consideration but that can be taken into account with conventional techniques. The DCO of  FIG. 4  is replaced in this Figure by the VCO  26 . 
         [0039]    What can be observed in  FIG. 5  is that for single bit phase counters the subtraction operation performed by unit  24  becomes very close to an EXOR operation because the unit  24  only produces an output when the count stored in the counters  22   a ,  22   b  is different. This shows the logical connection to more traditional PLL&#39;s. 
         [0040]    In the block diagram of  FIG. 5 , all edges on the input feed into the loop in the time domain so that their precise time information does not get lost. The bandwidth of the PLL can remain quite large, which is attractive. 
         [0041]    The information after the counters now spans a larger phase than is the case in the embodiment of  FIG. 1 . Such a phase detector would generally be designed to handle a phase range between −π and +π, while in circuit of  FIG. 5 , the range would need to be ±many π. If this information is present in the voltage domain, as would be the case for a voltage controlled oscillator there would more than two or three levels would be required. A phase detector may supply for instance 0, 1 and tri-state as applicable levels, which makes the step size for each phase change on the inputs smaller. However, that information is now in the amplitude domain, rather than in the time domain anymore. 
         [0042]    In the amplitude domain it is relatively simple to perform modulo (MOD) and multiplication operations, which in the time domain would require large blocks that are difficult to design. 
         [0043]      FIG. 6  shows an embodiment of the invention wherein a scaling stage  28   a ,  28   b  is added and the dividers are removed. As in  FIG. 5 , the input is unsampled. The output of the subtraction unit  24  is applied to a digital-to-analog converter  30 , which in turn is connected to filter, whose output is applied to VCO or CCO  26 . 
         [0044]    The M/N scaling stage  28   a  divides the phase angle by N and multiplies it by M. Thus the phase line after the scaling appears to run at a frequency that is M/N times the input frequency. The feedback frequency must be comparable with the scaled frequency, which was the sought after output frequency. The feedback frequency will be scaled with unity. 
         [0045]    The multipliers generally have a limited accuracy, but that implicit problem can actually be well contained. The range from the input counters  22   a ,  22   b  is limited by the MOD operators. Thus the multiplications in the scaling units  28   a ,  28   b  will impart only a limited error, which can be designed to sufficiently low levels which are satisfactory for the overall performance of the PLL. 
         [0046]    In the longer term the two MOD operators (N and M) will force the appropriate frequency behavior. For every N input cycles the output will have M/N*N=M output cycles, which can be handled by the M MOD operator. Thus, long term the errors in scaling cannot accumulate as the counters will incur the same quantities of actual MOD activations if the input and the output are correctly locked. 
         [0047]    The above embodiment thus provides a robust method to generate an output frequency which is M/N*input frequency without the use of time dividers. Off course there, many variants of this method are possible. 
         [0048]    The scaling factors can be changed around. For instance could the scaling factors be M for the input and N for the feedback, which could give implementation advantages. In general the input scaling could be M 1 /N 1 , and the feedback scaling N 2 /M 2 . This would lead to a total frequency transfer of M 1 *M 2 /N 1 *N 2 . A block diagram of such an arrangement is shown in  FIG. 7 . 
         [0049]    In another embodiment, the MOD operators with the same integer factor can be changed so that the phase range is changed (expanded). 
         [0050]    In another embodiment the subtraction unit is arranged downstream of a pair of DACs  30   a ,  30   b , as shown in  FIG. 8 . When the DACs are located upstream of the subtraction unit  24 , two DAC&#39;s are required as shown. The scaling can be in the analog or the digital domain. In  FIG. 8 , the scaling is in the analog domain, whereas in  FIG. 9  shows an embodiment wherein the scaling occurs in the digital domain. In this case, the MOD units  28   a ,  28   b  are located downstream of the DACs  30   a ,  30   b.    
         [0051]    In the embodiments described above, the input blocks have been sequenced slightly different. It is also possible to include part of the filtering operation, such as decimation, in the input stage.  FIG. 10  illustrates such an embodiment, where decimators  32   a  and  32   b  are included upstream of the MOD units  30   a ,  30   b.    
         [0052]    The introduction of a decimator into the block diagram useful to implement part of the PLL function in software. Software typically cannot operate at the speed of the incoming edges of the input signal, so that decimation down to a lower signal rate is necessary. The embodiment shown in  FIG. 10  is simple to implement in software, wherein the part downstream of the decimation is implemented in software. 
         [0053]    It is also possible to expand this operation by including the MOD operation in software. In this approach the hardware that counts the edges on the inputs can have almost any MOD operator (there will be an implicit one anyhow as the hardware will have limited size of memory), and the software can performs its own MOD operator by using a differentiator plus an integrator, the latter inclusive of the MOD operator. This approach leads to the implementation shown in  FIG. 11 . In this embodiment, the input counters  28   a ,  28   b  are simple up counters. Mod units  40   a ,  40   b  are placed between differentiators  42   a ,  42   b.    
         [0054]    The information in the DCO is represented by the output signal, sampled and processed again. Actually that information is already present in the DCO, and the presence of a physical representation of the information somewhere in the feedback is actually not required. Another embodiment of the invention can be as illustrated in  FIG. 12 . In this embodiment, a single up counter  28  feeds a single decimator  22 . The sampling unit  22 , the up counter  28  and the decimator  32  are implemented in hardware. The remaining part of the PLL is implemented in software. 
         [0055]    The feedback is replaced by a software path, which means that the unit requires less hardware that uses area and power, and that the hardware is as simple as possible, so for instance without a programmable MOD operator. This makes this embodiment highly attractive for many applications. 
         [0056]    The embodiment in  FIG. 12  is missing a physical output signal. However, it is well how to convert a software DCO running at a certain frequency into a hardware generated signal. This actually can be performed with frequency conversions so that it is possible to generate several signals from a single software PLL. It is apparent that in this embodiment the actual PLL loop is contained completely in software. This implies that the software also supplies full flexibility on the actual transfer curves. 
         [0057]    The MOD operators make the phase signal change from a straight sloping line into a sawtooth signal. The subtraction of the scaled sawtooth shaped signals may create overflow problems as demonstrated in  FIG. 13 . A first phase difference of the scaled phases is found ts A, which in size is Φ 1 −Φ 2 . If the same operation is performed at B that subtraction would become very negative, as the Φ 1  line is already over the ‘MOD top’ while the Φ 2  line is not. The mathematical solution to this problem is to switch at B to Φ 1 −Φ 2 +modval. In software or digital solutions that is a numerical operation; in an analog environment it involves the use of extra current.