Abstract:
A method for contactless charging of the battery of an electric automobile by magnetic induction using a transmitter coil of a charging device and a receiver coil of the vehicle, the method including: controlling a power supply of a converter, at terminals of which the transmitter coil is connected, according to a variable frequency; measuring, in an analog circuit, a value of a current and of a voltage at the terminals of the transmission coil; calculating a phase shift between the current and the voltage; converting the phase shift into a digital value; and locking the variable frequency of the converter to the phase-shift value by digital processing.

Description:
BACKGROUND 
     The present invention relates to the contactless charging of a battery of an electric or hybrid automotive vehicle. 
     Charging is performed by magnetic induction: in a location called a “charging zone”, a current is made to flow in a ground circuit possessing an emitting coil—or primary, thereby providing the power to a receiving coil—or secondary, of an electric or hybrid automotive vehicle, hereinafter simply referred to as a vehicle. 
     The phenomenon of magnetic induction takes place only if the primary and secondary coils are sufficiently close together, and the power transmitted depends in part on the resonance of the ground circuit. Although the vehicle is stationary when it is being charged, the frequency of the current flowing in the circuit must be adapted as a function of the position of the secondary with respect to the primary, therefore of the position of the vehicle in the charging zone with respect to the (stationary) primary. This is in order to achieve resonance of the system. 
     Hence more precisely, according to a first of its subjects the invention relates to a method of contactless charging of a battery of an electric automotive vehicle by magnetic induction between an emitting resonant circuit comprising an emitting coil of a charging device and a receiving resonant circuit of the vehicle comprising a receiving coil, the vehicle being positioned above the emitting coil, so as to be able to ensure good magnetic coupling between the emitting coil and the receiving coil, the method comprising steps consisting in:
         commanding the electrical power supply together with the setpoints of an inverter across the terminals of which is linked the emitting coil according to a variable frequency,   measuring in an analog circuit the value of the current and of the voltage across the terminals of the emitting coil, and   computing the phase shift between the current and the voltage.       

     Such a method is known to the person skilled in the art, especially through the example given thereof in the prior art document EP2317624 which is aimed especially at comparing the phases between the voltage and the current so as to drive the excitation frequency, in a circuit which comprises a “phase comparator” module, on the basis of logical signals which are images of the sign of the current and of the voltage, so as to generate a signal whose variable amplitude causes a variation of the excitation frequency by a VCO module, that is to say a voltage controlled oscillator which generates a signal whose frequency depends on the input voltage. 
     Other solutions consist in causing the frequency of the charging circuit to vary as a function of the power received at the secondary; but in this case, the battery of the vehicle is liable to refuse too great a transfer of power. 
     However, such solutions are complex and expensive to implement. 
     BRIEF SUMMARY 
     The aim of the present invention is to remedy these drawbacks by proposing a simple and essentially digital solution. 
     With this objective in view, the method according to the invention, moreover in accordance with the preamble cited hereinabove, is essentially characterized in that it furthermore comprises steps consisting in:
         converting the phase shift into a numerical value, and   slaving by a digital processing the frequency of the switching setpoints dispatched to the inverter to the value of the phase shift.       

     By virtue of the invention, the detection of the phase difference between voltage and current is very simple at the hardware level, it minimizes the number of hardware electronic components, and therefore the cost of implementation, an essential part being done digitally. Preferably the phase shift value corresponds to the resonance of the system. 
     By virtue of these characteristics, the adjustment of performance in the digital part is easy and parametrizable. 
     By virtue of the invention, it is possible to seek resonance at much reduced transferred power, this being useful since a vehicle battery is liable to refuse too great a transfer of power. 
     In one embodiment, steps are envisaged consisting in
         establishing the absolute value of the phase shift,   computing the derivative of the absolute value of the phase shift,   computing the sign of the derivative, and   estimating the value of the real phase shift as a function of the absolute value of the phase shift, of the derivative, and of its sign.       

     By virtue of this characteristic, it is possible by simple digital processing to know whether the frequency of the circuit is greater or less than the resonant frequency. 
     In one embodiment, a step is envisaged consisting in slaving the variable frequency to a phase shift value below a predetermined value. 
     This makes it possible to minimize the phase shift between the phase of the current and the phase of the voltage. This characteristic affords robustness of the system through the use of a closed loop on the phase. 
     By virtue of the invention, the frequency slaving is implemented by a software loop, thereby allowing complete flexibility of fine tuning of the slaving (frequency completely variable as a function of need). 
     Advantageously, the slaving is implemented by a proportional integral regulator the value of whose proportional gain and/or the value of whose integral gain are chosen so as to optimize the rate of convergence of the variable frequency to the predetermined value. 
     In one embodiment, at least one step of filtering the absolute value of the phase shift is envisaged. 
     In one embodiment, steps are envisaged consisting in
         comparing the derivative of the absolute value of the phase shift with a high threshold value, and with a low threshold value,   emitting a signal representative of the sign of the signal of phase shift between the current and the voltage,       

     in which
         if the derivative of the absolute value of the phase shift is greater than the high threshold value then the sign of the signal of phase shift between the current and the voltage is considered to be positive, and   if the derivative of the absolute value of the phase shift is less than the low threshold value then the sign of the signal of phase shift between the current and the voltage is considered to be negative.       

     According to another of its subjects, the invention relates to a computer program, comprising program code instructions for the execution of the steps of the method according to the invention when said program is executed on a computer. 
     According to the invention, the formulation of the phase is constructed in part in analog, and in part in digital (by software); this makes it possible to reduce the number of analog components, therefore to reduce the cost and to increase the reliability of the system, whereas a purely analog solution is very complex to implement. 
     Finally, the invention also relates to a system for contactless charging by magnetic induction of a battery of an electric automotive vehicle, a receiving resonant circuit of which comprises a receiving coil, 
     the system comprising:
         an emitting resonant circuit comprising an emitting coil,   an inverter across the terminals of which is linked the emitting coil,   an analog and digital motherboard on which are disposed an analog circuit and a control board, the current and the voltage across the terminals of the emitting coil being measured and processed by the analog circuit which computes the absolute value (ABS_PH_ANA) of the phase shift between the current and the voltage, the phase shift signal (ABS_PH_ANA) arising from the analog circuit being injected as input into the digital control board which emits as output a frequency setpoint (N_PWM) to the inverter, and   an analog-digital converter for converting the phase shift into a numerical value,       

     the control board being configured so as to slave by a digital processing the frequency of the switching setpoints dispatched to the inverter to the value of the phase shift, so as to be able to ensure magnetic coupling between the emitting coil and the receiving coil when the vehicle is positioned above the emitting coil. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Other characteristics and advantages of the present invention will be more clearly apparent on reading the following description given by way of illustrative and nonlimiting example and with reference to the appended figures in which: 
         FIG. 1  illustrates an overall diagram of a contactless charging of a vehicle, 
         FIG. 2  illustrates certain electronic details of the diagram of  FIG. 1 , 
         FIG. 3  illustrates an embodiment of a circuit for measuring the phase shift in absolute value between the voltage and the current, 
         FIG. 4  illustrates an embodiment of a part of the computer (processor) according to the invention, 
         FIG. 5  illustrates an embodiment of the block B 3  of  FIG. 4 , 
         FIG. 6  illustrates the evolution of the phase shift signal as a function of the variable frequency, 
         FIG. 7A  and  FIG. 7B  illustrate in a synchronous manner respectively the evolution of the resonant frequency and the evolution of the phase shift signal (signed and unsigned in absolute value) as a function of time in case of relative movement between the primary and the secondary, 
         FIG. 8  illustrates the evolution of the phase shift signal and its absolute value as a function of the variable frequency, 
         FIG. 9  illustrates an embodiment of the method according to the invention. 
     
    
    
     DETAILED DESCRIPTION 
       FIG. 1  illustrates the general diagram of a contactless charging of a vehicle, and  FIG. 2  illustrates certain electronic details of this general diagram, including a filter Filt. 
     An electrical source  1 , typically an electrical network, emits a sinusoidal current to a rectifier  2 . The rectifier  2  makes it possible to supply an inverter whose frequency is adjustable. The inverter  3  supplies a resonant charging circuit  10 , termed an LC circuit, comprising a charging coil  11  also called the primary or emitting coil. 
     The primary  11  is able to charge the battery of a vehicle  30  equipped with a resonant receiving circuit  20 , comprising a receiving coil  21  also called the secondary coil or simply the “secondary”. 
     The charging circuit  10  and the receiving circuit are configured so as to resonate at the same resonant frequency. 
     Now, the resonant frequency depends on the relative position of the primary and the secondary. 
     To drive the value of the resonant frequency, it is proposed to take action only at the level of the charging, that is to say not at the level of the geographical position of the vehicle. Accordingly, action is taken at the level of the inverter  3  whose frequency is driven by a regulator. 
     The current U and the voltage I across the terminals of the primary  11  are measured and processed by an analog circuit  41  which computes the absolute value ABS_PH_ANA of the phase shift between the current and the voltage, hereinafter referred to as the phase shift signal or “phase shift”. 
     The phase shift signal ABS_PH_ANA arising from the analog circuit  41  is injected as input into a digital control board  42  which emits as output a frequency setpoint which is an image of N_PWM to the inverter  3 . 
     In this instance, the analog circuit  41  and the control board  42  are disposed on an analog and digital motherboard  40 . 
     In a resonant circuit, the voltage and the current are in phase at resonance. The aim here is to perform a transfer of power between the primary  11  and the secondary  21  at resonance. 
       FIG. 2  presents more precisely the various power stages, and  FIG. 1  represents in detail especially the motherboard or block  40  which corresponds to the control. 
     As illustrated in  FIG. 6 , the phase difference between the voltage U and the current I in the charging circuit  10  is almost zero at resonance. The phase difference is negative before the resonant frequency, close to zero at resonance and positive beyond resonance (the phenomenon being reversed if the phase difference is established between the current I and the voltage U in the charging circuit). Thus, the absolute value of a signal corresponding to the phase difference between the voltage and the current in the charging circuit—termed the signal of absolute value—passes through an almost zero inflection point at resonance. 
     In the case illustrated in  FIG. 6 , which represents the absolute value of the phase shift as a function of frequency, the time derivative of the phase shift signal ABS_PH_ANA is therefore negative before the resonant frequency, and positive beyond. There is a correspondence between the sign of the phase difference and the sign of the derivative of the phase shift signal ABS_PH_ANA. 
       FIG. 8  represents the real value—that is to say with the sign—of the phase shift as a function of frequency. 
     To drive the establishment of the frequency of the inverter  3 , an analog processing is envisaged which makes it possible to provide the value of the phase shift ABS_PH_ANA, and a digital processing which makes it possible to provide an estimation of the sign of the phase difference, denoted PH_EST. 
     As represented in  FIG. 9 , at the analog level, the measurement  100  of the voltage U is compared with the measurement of the current I so as to compute  110  the value of the phase shift ABS_PH_ANA between these two quantities. For this purpose, a circuit  41  as illustrated in  FIG. 3  can be implemented. 
     In  FIG. 3 , U 1  and U 2  are comparators of the measurement of the voltage U (gating pulse signal) and the measurement of the intensity I (sinusoidal signal). Output from these comparators are positive gating pulse signals phase-shifted with respect to one another and input to an exclusive OR gate whose output is filtered, so as to recover the absolute value of the phase shift. 
     The (analog) phase shift signal ABS_PH_ANA is directed to an analog/digital converter at the input of a digital control board  42 , so as to convert  120  the phase shift into a numerical value. 
     At the digital level, a digital processing makes it possible to slave  130  the variable frequency of the inverter to the value of the phase shift. For this purpose, the control board  42  comprises a computer, and optionally the analog/digital converter mentioned hereinabove. 
     A (boolean) initialization signal init is directed to another input of the computer. The value of the initialization signal init indicates the order to perform the power transfer or not. This makes it possible to command  140  the electrical power supply of the inverter  3  across the terminals of which is linked the emitting coil  11  according to a variable frequency, which is the image of N_PWM. 
     For this purpose, the computer comprises a regulator, in this instance a pulse width modulation PWM regulator, whose output signal N_PWM corresponds directly to the chopping period of the voltage inverter  3 . The value of the signal N_PWM corresponds to a PWM register value and the chopping period T of the inverter  3  is related to the value of the signal N_PWM by the relation:
 
 T=N _PWM/10^8
 
     This formula being related to the clock cycles of the control board  2  (with a clock frequency equal to 100 MHz). 
     An embodiment of a part at least of the computer is illustrated in  FIG. 4 . It comprises four blocks B 1  to B 4 . 
     The blocks B 1 , B 2  and B 3  make it possible to construct, on the basis of the analog phase shift ABS_PH_ANA, the signal PH_EST representative of the sign of the phase shift. 
     The block B 1  is a filter which effects a filtering function, in this instance a low-pass filter of order  1 . The adjustment parameter (cutoff angular frequency) is fc_w, a typical value of which is for example 628 rad/s. The function of the block B 1  is to suppress the measurement noise arising from the analog signal ABS_PH_ANA. 
     The block B 2  is a differentiator, in this instance a differentiator filter, which carries out a differentiation function. Preferably, it also effects another filter of order  1  whose adjustment parameter is fc_w, a typical value of which is for example also 628 rad/s. As output from the block B 2  is generated a signal Der_ABS_PH, which is the image of the (filtered) derivative of the input signal ABS_PH_ANA. 
     The block B 3  formulates the signal sgn_PH representative of the sign of the derivative Der_ABS_PH. An exemplary embodiment of the block  3  is illustrated in  FIG. 5 . 
     The signal Der_ABS_PH arising from the block B 2  is compared with a first threshold value denoted Threshold_H: if the value of the derivative signal Der_ABS_PH is greater than this first threshold, then it is considered that the original signal (that is to say of the non-absolute-value phase shift signal) is positive and the passage to a positive value is detected; the signal DETECTION_PLUS_ 1  arising from the comparison equals 1. 
     The signal Der_ABS_PH arising from the block B 2  is also compared with a second threshold value denoted Threshold_L: if the value of the derivative signal Der_ABS_PH is less than this second threshold, then it is considered that the original signal is negative and the passage to a negative value is detected; the signal DETECTION_MINUS_ 1  arising from the comparison equals 1. 
     Threshold_H and Threshold_L constitute two adjustable parameters. Typical values are 10 and −10 (one of the thresholds is positive, the other is negative); (too low a value for Threshold_B risks falsifying the detections, and too high a value for Threshold_H risks not detecting changes of sign). 
     The signal DETECTION_MINUS_ 1  is multiplied by a predetermined value, in this instance the value 2, the result of which is added to the signal DETECTION_PLUS_ 1  to form the signal PATH_OUTPUT according to the following logic: 
     PATH_OUTPUT=0 if no detection of change of sign, 
     PATH_OUTPUT=1 if DETECTION_PLUS_ 1 =1, 
     PATH_OUTPUT=2 if DETECTION_MINUS_ 1 =1. 
     The output signal sgn_PH representative of the sign of the phase shift signal is thereafter easily constructed with the aid of a multiport switch for example according to the following logic: 
     If PATH_OUTPUT=0 then the sign computed during the last call of the block is retained, 
     PATH_OUTPUT=1, then sgn_PH=1, 
     PATH_OUTPUT=2, then sgn_PH=−1. 
     The output signal sgn_PH can also be looped as input via a block 1/Z “unit delay” ( FIG. 5 ) which makes it possible to recover the last value of the signal sgn_PH. In the case where no change has been detected, the previous value is then retained. 
     The output signal sgn_PH representative of the sign of the phase shift signal is multiplied with the phase shift signal ABS_PH_ANA to give the estimation signal PH_EST on one of the inputs of the block  4 . 
     The block  4  effects closed-loop regulation; for example a conventional regulator of the P-I (Proportional Integral) type. 
     In one embodiment, the principle of the slaving is to start from a low initial frequency, determined by the initial value, which corresponds for example to a typical initial value T_init equal to 9000. This value is dimensionally equivalent to a time, to return to seconds it is divided by 10^8 with a clock frequency equal to 100 MHz. 
     This initial value T_init is parametrizable. 
     Thus if the initial value equals 9000, the initial frequency of the inverter  3  then equals:
 
Frequency=10^8/9000=11 111 Hz.
 
     When the regulator is activated, the command of the frequency of the inverter  3  is fixed by the value of the signal N_PWM according to the relation:
 
 N _PWM= T _init− U _Corr
 
     U_Corr being the output of the P-I regulator (negative at the outset). 
     Thus the value of the signal N_PWM will decrease progressively and the frequency of the power supply will increase progressively until the desired resonance value. 
     The adjustment parameters (not represented in the figure) of the P-I corrector are:
         the proportional gain Kp,   the integral gain Ki.       

     These two parameters make it possible to adjust the rate of convergence of the variable frequency to resonance. 
     The first input of the block B 4  is a phase shift setpoint value CONS_PH that it is desired to achieve. On account of the various imperfections of the analog processing system making it possible to construct ABS_PH_ANA, the real phase shift signal never descends completely to zero, as indicated in  FIG. 6 . Therefore the setpoint cannot be fixed at 0 exactly. A typical value is 20° for example. 
     The second input corresponds to the quantity to be regulated, this being the reconstructed phase shift with its sign:
 
PH_ES=ABS_PH_ANA×sgn_PH
 
     The third input corresponds to the signal init (boolean signal indicating an initialization state); when the signal equals 1 the signal signifies a standby state for which one does not seek to transfer power, and when the output of the PI controller is equal to the value fixed by the fourth input, therefore here a zero value. 
     When the signal init equals 0, the output of the PI controller will decrease progressively (since the initial value is of the order of) −90°; the corrector will thus progressively decrease N_PWM starting from the value fixed by T_init, until the desired value (the value close to resonance). 
     The fourth input corresponds to an initial value VAL_INITIAL in this instance fixed at 0. No setpoint is dispatched as long as the charging order has not been given, that is to say the signal “init” has not passed to 0. 
       FIG. 7A  and  FIG. 7B  illustrate in a synchronous manner respectively the evolution of the resonant frequency and the evolution of the phase shift signal as a function of time, during a trial in which the receiver (vehicle) and the charging undergo several relative movements, which creates disturbances in the power transfer. 
     The primary is initially right opposite the secondary; the slaving imposes an initial frequency F 0  (in this instance about 21 kHz). 
     The secondary is firstly moved in one direction a first time. The frequency is decreased, progressively stabilized at a frequency F 1  (in this instance about 20.5 kHz). This movement temporarily increases the phase shift ( FIG. 7B ), the latter thereafter stabilizing around its setpoint value CONS_PH. 
     The secondary is then moved in the same direction a second time. The frequency is again decreased, progressively stabilized at a frequency F 2  (in this instance about 20 kHz). This movement again temporarily increases the phase shift ( FIG. 7B ), the latter thereafter stabilizing around its setpoint value CONS. 
     The secondary is then moved in the opposite direction to bring it back to the initial position. The slaving brings the frequency back to a value close to the initial value F 0  and this movement temporarily decreases the phase shift ( FIG. 7B ), the latter thereafter stabilizing around its setpoint value CONS_PH. 
     Note that the slaving has been adjusted here fairly sluggishly, but that it is possible to significantly accelerate the speed of the slaving.