Abstract:
A method and apparatus for a data access arrangement (DAA) which includes a line modulator containing capacitive elements to increase system stability. The invention provides improved stability during system startup and normal system operation. The line modulator is capable of adjusting the AC modulation and the DC termination presented to the telephone line. Capacitive elements are added to the modulator to provide enhanced system stability. The method includes drawing power from the telephone line, modulating the telephone line, sensing a level of distortion through the line modulator, feeding the sensed level of distortion to the line modulator, and using capacitive circuits to provide additional system stability.

Description:
FIELD OF THE INVENTION 
     This invention relates to a telephone line interface for data access arrangements (DAA). Specifically, it relates to a line powered DAA with enhanced stability. 
     BACKGROUND OF THE INVENTION 
     The telephone lines to a residence in the United States and elsewhere can have common mode voltages of over 100V, and the FCC requires the telephone lines to be isolated from any electric main powered device (such as a PC) connected to the telephone lines (through a modem for example) to prevent damage to the telephone network. 47 CFR 68.302,4 (Oct. 1, 1997 Edition). A data access arrangement (DAA) is specified by the FCC to isolate the telephone lines from electric main powered devices, such as illustrated in FIG.  4 . Since a voice band modem signal is limited to the 50 to 4000 Hz band, a DAA can be constructed using a transformer which operates as a bandpass filter to isolate the electric main powered device from the telephone lines. 
     A smaller size and potentially lower cost solution uses active circuits to communicate with the central telephone office and various modulation techniques to couple the DAA through small capacitors to a device, such as a PC. 
     FIG. 5 shows a known line powered telephone line interface circuit for modulating a data signal onto a telephone line using active circuits. The circuit is disclosed and described fully in U.S. patent application Ser. No. 09/028,061 filed on Feb. 26, 1998, entitled “Low Noise Line Powered DAA With Feedback,” assigned to the same assignee as the present application, and incorporated herein by reference. 
     The main function of the circuit is to take the incoming current I LINE  supplied by the telephone company and modulate it with a data signal developed by processing a differential data signal source V D  with a line modulator so as to place the data signal on the telephone line. The circuit uses transistor Q 1  as a line modulator, and contains a shunt regulator in series with the line modulator Q 1 . A sense resistor R S  is placed in series between the line modulator Q 1  and the shunt regulator to monitor the current through the shunt regulator. 
     The circuit depicted in FIG. 5 works by monitoring the current I S  through sense resistor R S  with a feedback loop around the amplifier A. Resistors R T  and R B  sense the differential voltage across R S . By setting R T =R B , the current through R T  and R B  will accurately model the current through R S . The desired signal to be modulated is introduced by a differential signal source V D . The differential signal is created by adding signal V D /2 to common mode voltage, V CM , to create V P , and subtracting V D /2 from V CM  to create V N . This differential signal then drives the input resistors R IP  and R IN  to provide a differential input current signal. The generation of the differential current signal is well known in the art. The control amplifier operates to force the current through resistor R S  to equal the desired signal current by regulating transistor Q 2  to control the base of transistor Q 1 , which in turn regulates the current through the collector-emitter path of transistor Q 1  and thereby through resistor R S . In this circuit, the collector current of transistor Q 1  is controlled by the control amplifier A. 
     Ideally, the current through R S  would equal the current, I LINE , introduced to the system by the telephone company. However, this is not the case in actuality. The current from the telephone company is introduced to the system through the emitter of transistor Q 1  (hereinafter “I E1 ”). In the circuit depicted in FIG. 5, I E1  is equal to I LINE , the resistances of R T  and R B  are a couple hundred thousand ohms, and the resistance of R S  is 10-20 ohms. Because of the relatively high level of resistance of R T1  and R B1 , the current that flows through R T1  and R B1  can be neglected in the circuit analysis. As current flows through the circuit, I E1  is divided into the transistor Q 1  base current (hereinafter “I B1 ”) and the transistor Q 1  collector current (hereinafter “I C1 ”) The collector current I C1  through the resistor R S  is used by amplifier A in a feedback loop to modulate the desired signal onto I LINE . 
     FIG. 6 shows another known line powered telephone line interface circuit for modulating a data signal onto a telephone line using active circuits. The circuit is disclosed and described fully in U.S. patent application Ser. No. 09/280,473 filed on Mar. 30, 1999, entitled “Method and Apparatus for Decreasing Distortion in a Line Powered Modulator Circuit,” assigned to the same assignee as the present application, and is incorporated herein by reference. 
     As in the circuit described above in reference to FIG. 5, the main function of the circuit in FIG. 6 is to take the incoming current I LINE  supplied by the telephone company and modulate it with a data signal developed by processing a differential data source signal V D  with a line modulator so as to place the data signal on the telephone line. The circuit uses transistor Q 1  as a line modulator, and contains a shunt regulator in series with the line modulator Q 1 . A first sense resistor R S1  is placed in series between the line modulator Q 1  and the shunt regulator to monitor the current through the shunt regulator. In addition, a second sense resistor R S2  is added within the modulator to pick up “stray components” of line current I LINE  which are outside of the feedback path containing the first sense resistor R S1 , and incorporate the “stray components” into an additional feedback path around the amplifier A. 
     The circuit depicted in FIG. 6 works by monitoring the current through sense resistor R S1  and R S2  with feedback loops around the amplifier A. The method of sensing the current through R S1  and R S2 , and for generating the differential signal current is similar to the circuit setup described in reference to FIG.  5 . The control amplifier operates to force the sum of the current through resistors R S1  and R S2 , and thereby I LINE , to equal the desired signal current by regulating transistor Q 2  to control the base of transistor Q 1 , which in turn regulates the current through the source-emitter path of transistor Q 1 . 
     FIG. 7 shows another known line powered telephone line interface circuit designed in low voltage CMOS technology for modulating a data signal onto a telephone line using active circuits. The circuit is disclosed and described fully in U.S. patent application Ser. No. 09/407,444 filed on Sep. 29, 1999, entitled “Pre-Charging Line Modem Capacitors to Reduce DC Setup Time,” assigned to the same assignee as the present application, and is incorporated herein by reference. 
     As in the circuits described above in reference to FIG.  5  and FIG. 6, the main function of the circuit in FIG. 7 is to take the incoming current, I LINE , supplied by the telephone company and modulate it with a data signal developed by processing a differential data signal source, V D , with a line modulator so as to place the data signal on the telephone line. The circuit uses transistor Q 1  as a line modulator, and contains a shunt regulator in series with the line modulator Q 1 . A sense resistor R S  is placed in series between the line modulator Q 1  and the shunt regulator to monitor the current through the shunt regulator. In addition, resistors R 1  and R 2  and capacitors C 1  and C 2  are included to set the AC gain of the modulator. Because the circuit is used to control AC and DC characteristics, additional components are required to obtain desired AC and DC values. 
     To have a low enough frequency response for the full voice band (e.g., down to about 50 Hz and up to about 4 kHz), capacitors C 1  and C 2  need to be large enough so there is not undue frequency response distortion of the signal. However, larger capacitors take a longer time to charge up, i.e., to finish settling down. The settling time may be as large as 400 ms, for example. During the time that the capacitors are charging, the transient charging current is added to the desired DC current level, causing a large error in the DC line current that lasts longer than a typical setup time limit, i.e., approximately 20 ms. Thus, when the line modulator is powered up, the time constant of resistor-capacitor pairs R 1 , C 1  and R 2 , C 2  cause DC current errors that result in a delayed DC setup time. 
     The circuit in FIG. 7 overcomes startup difficulties by using two precharge amplifiers A 2  and A 3  and two switches S 1  and S 2 . Amplifiers A 2  and A 3  are unity gain amplifiers which are used to precharge capacitors C 1  and C 2 , respectively, during startup. Precharge amplifiers A 2  and A 3  are configured as voltage followers to equalize nodes  1  and A with each other, and nodes  2  and B with each other, respectively. They are enabled and disabled in accordance with an enable signal EN. The goal is to raise node A up to the voltage at node  1  as quickly as possible, and to raise node B up to the voltage at node  2  as quickly as possible, thereby precharging capacitors C 1  and C 2 . Because no DC current passes through resistors R 1  and R 2 , the voltages at nodes A and B should equalize to those of nodes  1  and  2  very quickly, due to the operation of precharge amplifiers A 2  and A 3 . Amplifiers A 2  and A 3  are operational when they are enabled by a high enable signal EN. A high EN signal is generated from the beginning of the power up phase to some time period which allows the capacitors to be sufficiently precharged. The precharge operation in enhanced by adding switches S 1  and S 2 , which are also controlled by the EN signal so that they are open during the precharge phase. Switches S 1  and S 2  are placed in series with resistors R 1  and R 2  to eliminate them from the circuit during the precharge phase. 
     Although the circuits depicted in FIG. 5, FIG. 6, and FIG. 7 depict line powered DAAs capable of modulating a data signal onto a telephone line, the stability of the systems require improvement in order to create DAAs that are capable of being used for a wide variety of applications. Instability is introduced by the individual components of the line powered DAAs operating as amplifiers and voltage level shifters. Inherent to amplifiers and voltage level shifters is the potential for phase changes, especially at high frequencies. The phase changes associated with the individual components can produce oscillations in the DAA, which leads to system instability. 
     SUMMARY 
     The present invention proposes a novel method and apparatus for increasing the stability of a line powered telephone line interface or data access arrangement (DAA). The present invention accomplishes the task of increasing stability by strategically placing capacitance in the line powered telephone line interface or DAA. In a preferred embodiment, the invention enhances the normal operation of known DAAs by inserting additional capacitance into the DAA. In addition to enhanced system stability during normal operation, the present invention provides improved system stability on startup. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a circuit diagram of a stable, low noise, line powered DAA in accordance with the present invention. 
     FIG. 2 is a circuit diagram of the amplifier depicted in FIG. 1 portraying additional capacitance in accordance with the present invention. 
     FIG. 3 is a circuit diagram of an alternative embodiment of a line powered DAA in accordance with the present invention. 
     FIG. 4 is a block diagram of a conventional interface between a telephone network and an electric main powered device in accordance with the prior art. 
     FIG. 5 is a circuit diagram of a known data access arrangement (DAA). 
     FIG. 6 is a circuit diagram of a known data access arrangement (DAA). 
     FIG. 7 is a circuit diagram of a known data access arrangement (DAA). 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     The present invention provides a line powered data access arrangement (DAA) device having improved stability over the prior art. The present invention is particularly useful for modern telephone modems (V.90 standard). 
     A circuit diagram  10  of a preferred embodiment of the present invention is illustrated in FIG.  1 . The differential signal source  12  functions by adding half of the desired signal voltage  31  to the common mode voltage  34  to create voltage signal  30 , and subtracting half of the signal voltage  31  from common mode voltage  34  to create voltage signal  32 . These differential signals  30  and  32  then drive the input resistors  26  and  28  to provide a differential signal input current into the amplifier  24  at the non-inverting input  27  and at the inverting input  29 , respectively. The generation of the differential signal currents can be made by other means which are well known in the prior art, and thus will not be further discussed. 
     The shunt regulator  16  provides power drawn from the telephone line to the line modulator circuit  14  as well as to other modem and/or data processing circuitry necessary to provide a DC termination and AC modulation of the telephone line. The shunt regulator  16  limits the voltage across system components which are in parallel with the shunt regulator  16  to voltage level V DDA . The shunt regulator  16  is especially important if the amplifier  24  and other circuitry is fabricated in low voltage CMOS technology that cannot withstand voltages greater than 5 volts (or other fabrication technologies with low voltage requirements). Since the voltage difference between the telephone line tip voltage  36  and the telephone line ring voltage  38  can range from 5 to over 100 volts, the DAA  10  could be destroyed in the absence of shunt regulator  16 . 
     The sense resistor  18  is used in a feedback loop by the control amplifier  24  located within line modulator circuit  14 . By monitoring the current through sense resistor  18  with a feedback loop, the amplifier  24  can compensate for distortion in the DAA  10 . Resistors  17 A and  19 A sense the differential voltage across sense resistor  18 . By setting the resistance of resistor  17 A equal to the resistance of resistor  19 A, the current through resistors  17 A and  19 A will accurately model the current through resistor  18 . In a preferred embodiment, the resistance of resistors  17 A and  19 A is several hundred thousand ohms, while the resistance of the sense resistor  18  is approximately 10-20 ohms. Because of the relatively large resistance of resistors  17 A and  19 A, the current through these resistors can be neglected in the circuit analysis. Ignoring the currents through resistors  17 A and  19 A, the current through resistor  18  is approximately equal to the current through the shunt regulator  16 , which is connected in series with sense resistor  18 , and the current into other circuits connected to V DDA . Resistor R 1  and resistor R 2  are used to bias line modulator transistor  20 A in a commonly known manner. 
     Ideally, the feedback path to amplifier  24  will contain all of the telephone line current I LINE  which is introduced to the DAA  10 . The placement of sense resistor  18  will pick up a majority of the desired current, however, some stray current will pass outside of the feedback path containing the sense resistor  18 . If more of I LINE  is desired in a feedback path to amplifier  24 , auxiliary sense resistors in auxiliary feedback paths can be placed in circuit  10  without departing from the spirit of the present invention. 
     Resistor  17 B and capacitor  17 C coupled in series across feedback path resistor  17 A and resistor  19 B and capacitor  19 C coupled in series across feedback path resistor  19 A are used to set the AC gain of DAA  10 . During normal operation, amplifiers  17 D and  19 D are not activated and, therefor, do not affect circuit performance. Also, during normal operation, switches  17 E and  19 E are effectively closed allowing current to flow through resistors  17 B and  19 B. 
     During normal operation, the control amplifier  24  senses the current through sense resistors  18  with a feedback loop and attempts to control the circuit in the following manner. Resistors  17 A and  19 A sense the differential voltage across sense resistor  18 . By setting resistor  17 A equal to resistor  19 A, the current through resistor  17 A into the non-inverting input  27  of control amplifier  24  and the current through resistor  19 A into the inverting input  29  of control amplifier  24  will accurately model the current through sense resistor  18 . This sum approximately models I LINE  and is the parameter to be controlled. The feedback action of the loop comprising sense resistor  18 , amplifier  24 , control transistor  22 A, and line modulator transistor  20 A adjusts the current through sense resistor  18  such that the current through resistor  17 A equals the current from the differential signal source  12  through resistor  28 , and the current through resistor  19 A equals the current from the differential signal source  12  through resistor  26 . 
     Bipolar Junction Transistors (BJT) are used to describe the operation of line modulator transistor  20 A and control transistor  22 A. However, other types of transistors may be used, such as Field Effect Transistors (FET). For this reason the bases of transistors  20 A and  22 A may also be referred to as the control terminals, and the collector-emitter pairs of transistors  20 A and  22 A may be referred to as the current flow terminals of transistor  20 A and  22 A, respectively. 
     In accordance with a preferred embodiment of the present invention, as depicted in FIG. 1, the output of amplifier  24  is electrically connected to the emitter of control transistor  22 A, the collector of control transistor  22 A is electrically connected to the base of line modulator transistor  20 A, and the base of control transistor  22 A is electrically connected to the collector of line modulator transistor  20 A through sense resistor  18 . In this configuration, the sense resistor current I S  through the sense resistor  18  is equal to the line modulator transistor  20 A collector current I C1 . The total line current I LINE  consists of the transistor  20 A emitter current I E1  and the current through resistor R 1 . The modulator transistor  20 A collector current I C1  is nearly a linear function of its base current I B1  since the BETA of a transistor tends to be a weak function of the bias current. The feedback loop around sense resistor  18  controls I C1  to be a low distortion copy of the input signal. The current through resistor R 1  changes only a small amount because of the exponential relationship between the base-emitter voltage of transistor  20 A and the modulator transistor  20 A collector current I C1 . The effect of the current through resistor R 1  is further reduced if a transistor with a high BETA is used for modulator transistor  20 A so that the modulator  20 A base current I B1  will be small and a relatively small current is needed through resistor R 1 . 
     In DAA  10 , line modulator transistor  20 A is configured to function as an amplifier. Associated with line modulator transistor  20 A, configured as an amplifier, is the potential for a phase change at high frequencies. A phase change could cause DAA  10  to turn into an oscillator, resulting in instability of the DAA  10 . In order to control the phase change and enhance stability, resistor  20 B and capacitor  20 C are added in series between the base and collector of line modulator transistor  20 A. Resistor  20 B and capacitor  20 C form a resistance/capacitance circuit. Phase changes are most commonly associated with amplifiers operating at high frequencies. At high frequencies, capacitor  20 C behaves as a short, removing a portion of the transistor  20 A base current and causing the gain for high frequencies to be diminished in a controlled way, thereby controlling the phase and improving system stability. 
     Control transistor  22 A, in DAA  10 , is configured to function as a voltage level shifter. Associated with line modulator transistor  20 A, configured as a voltage level shifter, is the potential for a phase change at high frequencies. As discussed above, a phase change could result in instability of the DAA  10 . In order to enhance stability, resistor  22 B and capacitor  22 C are added in series between the emitter and collector of control transistor  22 A. Resistor  22 B and capacitor  22 C form a resistance/capacitance circuit. At high frequencies, capacitor  22 C behaves as a short, bypassing the voltage level shifter  22 A for high frequencies, thereby controlling the phase and improving system stability. 
     In order to further enhance stability, additional resistance and capacitance can be added to amplifier  24 . FIG. 2 depicts an exploded view of amplifier  24 . Except for the novel alterations discussed below, amplifier  24  is a conventional amplifier that is well known in the art. The amplifier  24  comprises a biasing portion  212 , an input stage  214 , an intermediate stage  216 , and an output stage  218 . As stated above, associated with an amplifier is the potential for a phase change at high frequencies. To increase system stability, resistor  204  and capacitor  206  are added in series between the output  200  and drain of field effect transistor (FET)  208 , of amplifier  24 . The gate of FET  208  is located at the non-inverting input of amplifier  24  and the drain of FET  208  is located at an input to intermediate stage  216 . Resistor  204  and capacitor  206  form a resistance/capacitance circuit. At high frequencies, capacitor  206  behaves as a short, causing the gain for high frequencies to be diminished in a controlled way, thereby controlling the phase and improving system stability. Resistor  220  and capacitor  222  are system components which add additional resistance and capacitance to the DAA  10  for stability during the precharge phase. Resistor  220  and capacitor  222  form a resistance/capacitance circuit. During normal operation, switch  202  is open. Therefore, resistor  220  and capacitor  222  do not affect circuit performance during normal operation. 
     Referring again to FIG. 1, during system precharge, amplifiers  17 D and  19 D are enabled. While enabled, amplifiers  17 D and  19 D charge capacitors  17 C and  19 C, respectively. Amplifiers  17 D and  19 D are unity gain amplifiers which alter the system stability of DAA  10  when in use during the precharging phase of DAA  10 . Amplifiers  17 D and  19 D are configured as voltage followers to equalize nodes  1  and A with each other and nodes  2  and B with each other, respectively. The goal is to raise node A up to the voltage of node  1  as quickly as possible, and to raise node B up to the voltage of node  2  as quickly as possible, thereby precharging capacitors  17 C and  19 C, respectively. In a preferred embodiment, amplifiers  17 D and  19 D are introduced to the circuit by applying an enable signal EN to the amplifiers  17 D and  19 D. 
     In the preferred embodiment, switches  17 E and  19 E are opened when enable signal EN is applied to the switches  17 E and  19 E. Opening switches  17 E and  19 E effectively removes the paths containing resistors  17 B and  19 B from the DAA  10  during system precharge. Removing the paths containing resistors  17 B and  19 B reduces the effects of the offset voltages of amplifiers  17 D and  19 D and the DC voltage drops in the precharge current paths. If not removed, these voltages would be effectively across resistors  17 B and  19 B, and would develop error currents into the summing nodes of amplifier  24 . 
     During system precharge, additional resistance and capacitance is needed to offset instability created while amplifiers  17 D and  19 D are charging their respective capacitors  17 C and  19 C. Referring to FIG. 2, resistor  220  and capacitor  222  are added to amplifier  24  by switch  202 . Switch  202  is closed during system precharge, thereby switching additional resistance and capacitance into DAA  10 , and opened during normal operation when additional resistance and capacitance is no longer needed. 
     FIG. 3, depicts a DAA  100  with a Darlington pair  300  substituted for the line modulator transistor  20 A in FIG.  1 . The Darlington pair  300  comprises transistor  300 A and transistor  300 B. The same stabilization technique as presented above can be used with the DAA  100  containing Darlington pair  300 . The Darlington pair  300  is substituted into the DAA  10  depicted in FIG. 1 by removing line modulator transistor  20 A; and connecting the emitter of transistor  300 A where the emitter of line modulator transistor  20 A was previously positioned, connecting the base of transistor  300 B where the base of line modulator transistor  20 A was previously positioned, and connecting the collectors of transistors  300 A and  300 B where the collector of line modulator transistor  20 A was previously positioned. Resistor R 1A  is added for biasing transistor  300 B. The resistance and capacitance technique used for stabilization as set forth in the discussion of FIG.  1  and FIG. 2 remains the same for the use with the Darlington pair  300 . The base of transistor  300 B is herein termed the base (control terminal) of Darlington pair  300 , and the emitter and collector of transistor  300 A are herein termed the emitter and collector (current flow terminals) of Darlington pair  300 . 
     Accordingly, the present invention provides a low noise DAA that is particularly useful for modern modems which require a high level of stability. 
     Having thus described a few particular embodiments of the invention, various alterations, modifications, and improvements will readily occur to those skilled in the art. Such alterations, modifications and improvements as are made obvious by this disclosure are intended to be part of this description though not expressly stated herein, and are intended to be within the spirit and scope of the invention. Accordingly, the foregoing description is by way of example only, and not limiting. The invention is limited only as defined in the following claims and equivalents thereto.