Abstract:
A method for analog-to-digital conversion of a chronologically changing, analog input signal includes the steps of forming an edited signal from an input signal by subjecting the input signal to a first transfer function and amplifying the result by a defined gain factor, sampling the input signal and the edited signal and subjecting the sampled signals to analog-to-digital conversion to obtain first and second digital data respectively corresponding to the edited signal and the input signal. The number of bits in each of the first and second digital data determines the dynamic range of the analog-to-digital conversion. The first digital data, corresponding to the input signal, are multiplied by a defined factor equal to the defined gain factor. If a transgression of the dynamic range in the analog-to-digital conversion has not occurred, the final analog-to-digital conversion result is calculated making use of the second digital data corresponding to the edited signal. If the dynamic range has been transgressed, the digital data corresponding to the edited signal are not used to determine the final conversion result. An apparatus for practicing the method is also disclosed.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention is directed to a method and apparatus for analog-to-digital conversion of a chronologically changing analog electrical input signal. 
     2. Related Application 
     The subject matter of this application is related to the subject matter of an application filed simultaneously with the present application and having Ser. No. 07/662,403, and assigned to the same assignee as the present application. 
     3. Description of the Prior Art 
     Analog-to-digital converters in the form of integrated circuits are currently available which operate, for example, according to the &#34;dual slope&#34; method, and permit analog-to-digital conversion of a chronologically changing analog electrical input signal with a resolution of up to 16 bits. The use of analog-to-digital converters for some purposes, such as electrocardiography (ECG) or electroencephalography (EEG), require that such converters be operated so that the smallest acquirable voltage difference of the signal to be measured (which corresponds to the least significant bit (LSB) of the analog-to-digital conversion) is on the order of magnitude of a few microvolts. Under such conditions, known analog-to-digital converters have an extremely limited dynamic range for the signal to be measured. This results in the analog-to-digital converter being very quickly overdriven by the signal to be measured. This is particularly undesirable in the case of electrocardiography, wherein registration of an electrocardiogram should be possible immediately after a defibrillation. To effect defibrillation, the heart of a patient is charged with pulses having relatively high voltage and current intensity. It is conceivable to alleviate the problem of overdriving the analog-to-digital converter by using analog-to-digital converters having higher resolution (number of bits). If the analog-to-digital converter is to be executed as an integrated circuit, the manufacture of such an integrated converter having a sufficiently high resolution is extremely expensive, and the alternative of using a discretely constructed analog-to-digital converter involves a considerable technological and financial outlay. 
     Such outlay can be at least partially avoided by using so-called DAC feedback. Using this known technique, the digital output signals of the analog-to-digital converter are supplied to an electronic calculating means, which drives a digital-to-analog converter connected to the electronic calculating means so that it generates an analog output signal, which is then subtracted in a differential amplifier from the signal to be measured. The output from the differential amplifier is supplied to the analog-to-digital converter. The digital-to-analog converter is driven by the electronic calculating means dependent on the momentary amplitude of the signal to be measured, so that the output of the differential amplifier does not exceed the dynamic range of the analog-to-digital converter. The digital data corresponding to the output signal of the differential amplifier are subsequently corrected in the electronic calculating means by adding the corresponding value of the output signal of the digital-to-analog converter thereto. Exact results are only obtained with this method, however, if the resolution (number of bits) of the digital-to-analog converter corresponds to the desired resolution of the analog-to-digital conversion. Problems similar to those described above in conjunction with analog-to-digital converters also occur in digital-to-analog converters, therefore digital-to-analog converters are usually employed having a resolution which is lower than the desired resolution of the analog-to-digital conversion. This results in imprecisions in the conversion. For example, the results of the analog-to-digital conversion contain discontinuities which are not present in the analog input signal. A further disadvantage of this known technique is the necessity of an additional digital-to-analog converter, which results in additional costs. 
     SUMMARY OF THE INVENTION 
     It is an object of the present invention to provide a method and apparatus for analog-to-digital conversion of a chronologically changing, analog electrical input signal which are technologically simple and economic, which have a large dynamic range, and which permit the measurement of extremely small voltage changes. 
     It is further object of the present invention to provide such a method and apparatus which are suitable for analog-to-digital conversion of electrocardiographic signals. 
     The above objects are achieved in accordance with the principles of the present invention wherein an edited signal is formed and digital data are obtained from the edited signal as well as from the input signal. The edited signal is formed by subjecting the input signal to a first transfer function, and amplifying the result by a defined gain factor. The input signal will therefore include signal portions which are not present in the edited signal due to the application of the transfer function. As a consequence of the amplification by the defined factor, a change in the result of the analog-to-digital conversion of the edited signal by one least significant bit (LSB) corresponds to a voltage change of the input signal which is equal to the difference in voltage corresponding to a LSB divided by the defined gain factor. If the result of the analog-to-digital conversion is calculated using the digital data corresponding to the edited signal, the measurable differences in the voltage, as a consequence of the amplification, are smaller than in the case of calculating the result of the analog-to-digital conversion solely on the basis of the digital data corresponding to the unamplified input signal. 
     With respect to the edited signal, however, a reduction of the dynamic range, corresponding to the defined gain factor, arises. If signal parts contained in the input signal which were removed from the edited signal due to the application of the transfer function are of interest, one can proceed in the simplest case by using the digital data obtained by analog-to-digital conversion of the edited signal as the final conversion result as long as the dynamic range of the converter used to conduct the analog-to-digital conversion is not transgressed. If this dynamic range is transgressed, the result of the analog-to-digital conversion of the input signal is used as the final conversion result. 
     The method therefore allows a large dynamic range to be realized, while simultaneously permitting extremely small changes in voltage to be measured in a technologically simple and economic manner, with a given number of bits of the resultant digital data. As a consequence of multiplication of the digital data corresponding to the input signal by a multiplication factor which is equal to the defined gain factor, the digital data corresponding to the input signal and the digital data corresponding to the edited signal are directly comparable, if it is assumed that the edited signal does not lack any signal parts of interest which are present in the input signal. 
     An embodiment of the invention which is of particular advantage for electrocardiography uses high-pass filtering of the input signal as the first transfer function. This insures that the edited signal correctly reproduces the dynamic part of the signal, even after defibrillation, without transgressions of the dynamic range, because those signal parts of the input signal which correspond to the tissue polarization caused by the defibrillation are filtered out. 
     In a further embodiment of the invention, the digital data corresponding to the input signal are subjected to a further transfer function, which is the complex complement of the first transfer function. Subjecting data to successive transfer functions, one of which is the complex complement of the other, is described in the book &#34;Theory and Application of Digital Signal Processing,&#34; Rabiner et al. As a consequence of subjecting the digital data corresponding to the input signal, which had been multiplied by the defined multiplication factor, to the further transfer function which is the complex complement of the first transfer function, the resulting digital data correspond to the difference between the chronological curves of the input signal and of the edited signal. The equivalent of an &#34;interpolation&#34; takes place between the bits which are separated from one another by a LSB in the data corresponding to the input signal, as a result of the digital filtering used to apply the transfer function. When the digital data obtained by the digital filtering of the input signal are added to the digital data corresponding to the edited signal in chronologically correct allocation, digital data having a number of bits which is higher than the number of bits of the analog-to-digital conversion are obtained as a sum, as a consequence of the interpolation achieved by the digital filtering. The number of bits is higher to an extent corresponding to the defined gain factor. In the case, for example, of a defined gain factor of 32, an increase of 5 in the number of bits is achieved, whereas a defined gain factor of 64 would yield an increase of 6 bits. As long as no transgressions of the dynamic range occur in the analog-to-digital conversion of the edited signal, digital data having a number of bits which is higher than the number of bits of the analog-to-digital conversion will be obtained. It is important that the digital data obtained in the manner set forth above cover those signal parts which were removed in the formation of the edited signal due to the application of the first transfer function. Digital data having a number of bits which is limited to the number of bits of the analog-to-digital conversion must be used, as previously described, only when a transgression of the dynamic range occurs in the analog-to-digital conversion of the edited signal. 
     This limitation may also be overcome in a further embodiment of the invention. Without taking the digital data corresponding to the edited signal into consideration, digital data having a number of bits higher than the number of bits of the analog-to-digital conversion are obtained in this embodiment as a result of analog-to-digital conversion of the input signal. The reason for this is again that an &#34;interpolation&#34; occurs between the data bits. The digital data corresponding signal are again multiplied by the defined multiplication factor and the product is subjected to an integrating transfer function by digital filtering. Again, an &#34;interpolation&#34; takes place between bits of the product data which are separated from one another by one LSB. The time constant of the integration is preferably selected so that the integration covers a plurality of sets of the digital data which corresponds to the defined gain factor. The digital data used as the data corresponding to the input signal have a number of bits which is 5 bits more than the number of bits of the analog-to-digital conversion given a gain factor of, for example, 32. The analog electrical signal for this purpose, and analog electrical signal is superimposed on the input signal before the input signal is subjected to the analog-to-digital conversion. The analog electrical signal may be a periodic signal, for example a sawtooth signal, or may be a non-periodic signal, such as noise. The only important criterion is that the superimposed analog signal have a statistically uniform amplitude distribution. 
     Without taking the digital data which correspond directly to the edited signal into consideration in calculating the conversion result, one obtains digital data as a result of the analog-to-digital conversion whose number of bits is higher than the number of bits in the analog-to-digital conversion. The reason for this is that again an &#34;interpolation&#34; ensues between digital data which are separated from one each other by one LSB, which correspond to the input signal and which are multiplied by the defined factor. This ensues by the application of the integrating transfer function, which is undertaken by digital filtering. The time constant of the integration is preferably selected so that the integration covers digital data corresponding to the defined gain factor. The digital data are then used as digital data corresponding to the input signal, but have a number of bits which is larger by 5 bits than the number of bits of the analog-to-digital conversion given, for example, a gain factor of 32. The analog signal, for example, may be periodic signal, for example a sawtooth signal, or of a non-periodic signal, for example noise. The only important factor is that the signal has a statistically uniform amplitude distribution. Because the method in this embodiment of the invention has somewhat less precision in comparison to the aforementioned embodiments, particularly in view of the dynamic part of the input signal, this embodiment should only be used under the circumstances wherein the unique advantages thereof are required. 
     An apparatus operating according to the above method is also disclosed herein. A method for calibrating the disclosed apparatus is also disclosed which permits the defined gain factor and the defined multiplication factor to be exactly matched to each other, and if necessary permits tuning of the second transfer function to the first transfer function to be undertaken to assure that the second transfer function in fact represents the complex complement of the first transfer function. 
    
    
     DESCRIPTION OF THE DRAWINGS 
     FIG. 1 a schematic block diagram of an apparatus for analog-to-digital conversion constructed in accordance with the principles of the present invention and operating in accordance with the inventive method. 
     FIGS. 2 through 5 respectively show analog circuits for illustrating different operating modes of the apparatus of FIG. 1. 
     FIG. 6 is a diagram for explaining a further operating mode of the apparatus of FIG. 1. 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     The apparatus shown in FIG. 1 may be a component, for example, of a means for processing and recording electrocardiographic signals. The apparatus of FIG. 1 includes a first 9:1 channel analog multiplexer 1. Usually nine ECG contacts are used (not shown, but including three contacts for the extremities and three contacts for the chest). These ECG contacts are respectively allocated to the inputs of the multiplexer 1. A second multiplexer 2, also a 9:1 channel analog multiplexer, is in parallel with the first multiplexer 1, for the purpose described below. The outputs of the two multiplexers 1 and 2 are connected to the inputs of a 2:1 channel analog multiplexer 3. The output signal of the multiplexer 3 is supplied to the input of an analog-to-digital converter 4 which is a 14-bit converter. The digital output data of the converter 4 proceed via a data bus 5 to an electronic calculating unit 6 in which they are further processed. A recorder 7, which may be an ECG strip plotter, is connected to an output of the calculating unit 6. 
     The signals corresponding to the individual ECG contacts do not proceed directly to the input of the first multiplexer 1. Instead, as shown in FIG. 1 with respect to one exemplary channel, the signals pass through an amplifier 8, which in the exemplary embodiment of FIG. 1 has a gain of 10. The output signal of the amplifier 8 is supplied to a sample and hold circuit 9. The output of the sample and hold circuit 9 is connected to an input of the first multiplexer 1. The output signal of the amplifier 8 is also supplied to a high-pass filter 10 of the first order, formed by a capacitor C and a resistor R. 
     The high-pass filter 10 applies a first transfer function T1 to the signal supplied to it, this first transfer function T1 being that of the high-pass filter 10 itself. The limit frequency of the high-pass filter 10 is selected sufficiently low to avoid suppression of diagnostically relevant, low-frequency parts of the input signal. 
     The output of the high-pass filter 10 proceeds to a second amplifier 11, having a gain factor V=32 in the exemplary embodiment. The output of the second amplifier 11 is supplied to a second sample and hold circuit 12, whose output is connected to one of the inputs of the second multiplexer 2, i.e., to that input corresponding to the input of the first multiplexer 1 which is connected to the first sample and hold circuit 9 associated with the respective channel. In the case of the exemplary embodiment of FIG. 1, the sample and hold circuits 9 and 12 are respectively connected to the second input (counted from the top) of the multiplexers 1 and 2. The sample and hold circuits 9 and 12 can be eliminated if it is assured that the repetition rate of the analog-to-digital conversions for each channel of the apparatus is clearly above the highest signal frequency of interest, or above the upper limit frequency of the supplied signal. 
     Even though it has been amplified in the amplifier 8, the signal supplied to the first sample and hold circuit 9 shall be referred to below as the input signal ES. The signal acquired from the input signal ES by removing the DC voltage components thereof with the high-pass filter 10 and by amplification by the defined gain factor V in the amplifier 11, and which is supplied to the second sample and hold circuit 12, shall be referred to as edited signal AS. 
     Via control lines, the multiplexers 1, 2 and 3 and the sample and hold circuits 9 and 12 are connected to the electronic calculating unit 6, from which they receive control signals. The multiplexers 1 and 2 are driven via a common control line so that the individual inputs thereof are respectively connected to the output in periodic succession, this operation being cyclically repeated. During the time interval for which the inputs of the multiplexers 1 and 2 associated with a defined ECG contact are connected to the respective outputs of the multiplexers 1 and 2, the calculating unit 6 supplies an inhibit signal to the respective sample and hold circuits 9 and 12 via a common control line. This assures that the voltage at the respective inputs of the multiplexers 1 and 2 does not change during this time interval. 
     The multiplexer 3 is controlled by the calculating unit 6 so that the multiplexer 3 connects the output of the multiplexer 1 to the input of the analog-to-digital converter 4 for one-half of the aforementioned time interval, and connects the output of the multiplexer 2 to the input of the analog-to-digital converter 4 for the other half of the time interval. The time interval is selected sufficiently long to enable the analog-to-digital converter 4 to undertake an analog-to-digital conversion during the respective halves of the time interval. The analog-to-digital converter 4 is connected to the calculating unit 6, which supplies the analog-to-digital converter 4 with a pulse at a suitable time. This pulse initiates the beginning of an analog-to-digital conversion. The analog-to-digital converter 4 generates an output signal supplied to the calculating unit 6, via the same line used to supply the control signal to the converter 4, when the converter 4 completes an analog-to-digital conversion. In the exemplary embodiment of FIG. 1, the calculating unit supplies the pulses which respectively initial the analog-to-digital conversions with repetition rate so that aliasing effects are avoided. The analog-to-digital converter 4 thus supplies the calculating unit 6 with a stream of digital data, which is obtained by the alternating, quasi-simultaneous analog-to-digital conversion of the input signal ES and of the edited signal AS of the ECG contacts corresponding to the individual channels. The digital data resulting from each conversion have the same number of bits, namely 14 bits. 
     As also shown in FIG. 1, a signal generator 13 is provided which generates a periodic signal PS having a statistically uniform amplitude distribution, for example a sawtooth signal. The periodic signal PS can be supplied to summing stages 16 and 17 via respective switches 14 and 15 which are actuatable by the calculating unit 6 via corresponding control lines. The summing stage 16 lies between the output of the sample and hold circuit 9 and the inputs of the multiplexer 1, and the summing stage 17 lies between the apparatus input and the input to the amplifier 8. There is thus the possibility of selectively overlaying the periodic signal PS on the samples of the input signal ES, or on the signals supplied to the amplifier 8. The frequency of the periodic signal PS is preferably selected so that its period duration corresponds to the time which is required for a plurality of analog-to-digital conversions of related samples of the input signal ES and of the edited signal AS, the plurality of related samples corresponding in number to the defined gain factor V. To achieve the required synchronization, the signal generator 13 is connected to the calculating unit 6 via a control line. 
     The electronic calculating unit 6 is constructed in a known manner and includes a central processing unit (CPU) 18, a read-out memory (ROM) 19 and a random access memory (RAM) 20, which are connected to each other via an address bus 21 and via a data bus 22. The ROM 19 and the RAM 20 serve the purpose of storing the operating system, the actual program and variable data. A crystal 23 is connected to the CPU 18, the crystal 23 generating clock signals which are required for operating the calculating unit 6 and from which the control signals for the multiplexers 1, 2, 3, the analog-to-digital converter 4, the sample and hold circuits 9 and 12, and the signal generator 13 are derived. These signals proceed via an input/output circuit (I/O port) 24 to these respective components. The input/output circuit 24 is connected both to the address bus 21 and the data bus 22. A data bus 5, via which the digital output signals of the analog-to-digital converter 4 proceed to the electronic calculating unit 6, is also connected to the input/output circuit 24. A keyboard 25, and a data display 26 and the aforementioned recording unit 7 are also connected thereto. 
     The digital data corresponding to the input signals ES and to the edited signals AS are processed in the calculating unit 6 so that, first, input signals ES can be converted with full exploitation of the dynamic range of the analog-to-digital converter 4 and, second, so that voltage changes of the input signal ES are smaller than that voltage change which would effect a change in the result of the analog-to-digital conversion by one LSB, given direct routing of the input signal ES to the input of the analog-to-digital converter 4, and without subsequent processing of the digital output data of the analog to digital converter 4. This shall be set forth below with reference to one channel in the embodiment FIG. 1, with reference to analog circuits respective shown in FIGS. 2 and 3. The analog circuits shown in FIGS. 2 and 3 illustrate the program executed by the calculating unit 6, insofar as it involves the processing of the data supplied to the calculating unit 6 from the analog-to-digital converter 4. FIGS. 2 and 3 are for explanatory purposes only, and should not be misinterpreted as meaning that an analog computer may be employed instead of the calculating unit 6 which is a digital computer. This would not be possible because the digital data supplied to the calculating unit 6 by the analog-to-digital converter 4 are already in digital form, and thus not capable of being processed with analog circuitry. 
     The electronic calculating unit 6 basically multiplies the digital data corresponding to the input signal ES, which are supplied by the analog-to-digital converter 4 and which have been obtained by the analog-to-digital conversion of the output signal of the sample and hold circuit 9, by a defined factor F. The factor F corresponds to the gain factor V with which the output signal of the high-pass filter 10 is amplified in order to acquire the edited signal AS. In the exemplary embodiment, the defined factor F is 32. This is indicated in FIG. 2 by an amplifier 27 having a gain of 32. 
     The electronic calculating unit 6 checks the digital data corresponding to the edited signal AS, which are supplied by the analog-to-digital converter 4 are obtained by the analog-to-digital conversion of the output signal of the sample and hold circuit 12, to determine whether a transgression of the dynamic range of the analog-to-digital converter 4 is present. This can e achieved in a simple manner by assuming that a transgression of the dynamic range has occurred if all bits of the digital data in a sample of the edited signal AS have the value of logic &#34;1&#34;. When the analog-to-digital converter 4 generates an output (overflow) indicting a transgression of the dynamic range, it is sufficient for the calculating unit 6 to monitor this signal. In FIG. 2, this type of checking is symbolized by showing the edited signal AS being supplied to a detector circuit 28, which generates an output signal when a transgression of the dynamic range occurs. 
     In the simplest case, the calculating unit 6 considers the digital data acquired from the analog-to-digital conversion of the edited signal AS to be the final result of the analog-to-digital conversion as long as there is no transgression of the dynamic range. If a transgression of the dynamic range has occurred, it considers the digital data acquired by an analog-to-digital conversion of the input signal ES, multiplied by the defined multiplication factor F, to be the final result of the analog-to-digital conversion for the duration of the transgression. This is symbolically indicated in FIG. 2 by showing the detector circuit 28 actuating an electronic switch 20 which, if there is no transgression of the dynamic range, assumes the position shown in FIG. 2, causing the edited signal AS to be supplied to the output A. If there is a transgression of the dynamic range, the output signal of the detector 28 switches the electronic switch 29 to its other state for the duration of the transgression, so that the output A is supplied with the amplified input signal ES. 
     The just-described first operating mode can be employed when it is primarily the dynamic part of the input signal ES, rather than the DC voltage part thereof, which is of interest. In this first operating mode there is the possibility, as long as there are no transgressions of the dynamic range by the edited signal AS, of measuring voltage changes of the dynamic part of the input signal ES which are smaller than the quotient of that voltage change which effects a modification of the result of the analog-to-digital conversion by one LSB (given a direct feed of the input signal ES to the input of the analog-to-digital converter 4), and the defined gain factor V. If a transgression of the dynamic range by the edited signal AS occurs, or if the DC voltage parts contained in the input signal ES are of interest, the data acquired by the analog-to-digital conversion of the product of the input signal ES and the defined factor F are accessed. The discontinuity which occurs as a result essentially corresponds only to the DC voltage part contained in the input signal ES, because the digital data corresponding to the input signal were multiplied by the factor F. 
     If the digital data corresponding to the input signal ES are to be free of DC voltage parts, there is the possibility of modifying the described operating mode so that the calculating unit 6 applies a second transfer function T2 to the digital data corresponding to the input signal ES. This may occur either before or after multiplication by the factor F. The second transfer function T2 is the transfer function of a high-pass filter, preferably that of the high-pass filter 10 which was employed in the formation of the edited signal AS. If the different resolution is not considered, the results of the analog-to-digital conversion are directly comparable to one another, independently of whether they were calculation using the digital data corresponding to the edited signal AS, or were calculated using the digital data corresponding to the input signal ES and without taking the digital data corresponding to the edited signal AS into consideration. The modified operating mode is symbolically illustrated in FIG. 3, which otherwise corresponds to FIG. 2, wherein the amplifier 27 is followed by a high-pass filter 31, formed by a capacitor C and a resistor R which can be bridged by the switch 30. 
     It is clear that the transfer functions T1 and T2 need not necessarily be the transfer functions of high-pass filters. If, for example, high-frequency signal parts are to be suppressed, the transfer functions may be the transfer functions of low-pass filters. It is also possible to use the transfer function of a notch filter, in which case defined noise frequencies can be filtered out. 
     In a second operating mode of the apparatus, the calculating unit 6 subjects the digital data which are the product of the input signal ES and the defined factor F to a third transfer function T3 by digital filtering. The third transfer function T3 is the complex complement of the first transfer function T1, which was used to eliminate DC voltage parts from the input signal ES in forming the edited signal AS. The calculating unit 6 adds the digital data obtained by the digital filtering to the digital data obtained by the analog-to-digital conversion of the edited signal AS, the addition taking place in chronologically correct relationship. The calculating unit 6 takes the result of this addition into consideration in formulating the result of the analog-to-digital conversion as long as there is no transgression of the dynamic range by the edited signal AS. If the edited signal AS transgresses the converter dynamic range, the calculating unit 6, as described above, considers the digital data corresponding to the product of the input signal ES and the factor F as the result of the analog-to-digital conversion, possibly after the application of the second transfer function T2 by digital filtering. 
     As is known, a mathematical function and its complex complement have the property that, when they are respectively multiplied by a third mathematical function, the sum of the products obtained in this manner corresponds to the third mathematical function. In the present case, this means that those signal parts which were lost when passing through the high-pass filter 10 are again &#34;attached&#34; to the edited signal AS as a result of this addition. This results in the DC components contained in the input signal ES being taken into consideration. Since an &#34;interpolation&#34; also occurs in the digital filtering between digital data corresponding to the input signal ES which are separated from each other by one LSB, the advantage is obtained that the digital data arising due to the aforementioned addition have a number of bits (given adequate precision of the calculating events undertaken in conjunction with the digital filtering) which is larger than the number of bits of the analog-to-digital converter 4. In the case of the exemplary embodiment, wherein the defined gain factor V and the defined factor F each have a value of 32, the calculating unit 6 executes the calculating events required for digital filtering with such precision that an increase of 5 in the number of bits (14) of the analog-to-digital conversion occurs, so that a total of 19 bits results. 
     In the exemplary embodiment, the input signal ES passes through the high-pass filter 10 of the first order for forming the edited signal AS. The complex complement of the transfer function T1 is used as the transfer function T2, which is realized by a low-pass filter of the first order whose limit frequency is identical to the limit frequency of the high-pass filter 10. 
     It should be noted that this will not be valid if a high-pass filter of a higher order is used instead of a high-pass filter of the first order. (If this is the case, correction elements must be added to the transfer function of the corresponding low-pass filter in order to obtain the complex complement). 
     The second operating mode is symbolically shown in FIG. 4, which otherwise corresponds to FIG. 2. In FIG. 4, the output of the amplifier 27 is supplied to a low-pass filter 32 of the first order, formed by the resistor R and the capacitor C. The values of the resistor R and the capacitor C correspond to those in the high-pass filter 10. The output of low-pass filter is added to the edited signal AS at the summing stage 33. As a consequence of those signal parts which were previously removed by the high-pass filter 10 being again added to the edited signal AS, a signal corresponding to the input signal ES amplified by a factor of 32 is thus available following the summing stage 33. As long as there is no transgression of the dynamic range by the edited signal AS, the results of the analog-to-digital conversion obtained by using the digital data corresponding to the edited signal AS offer an enhanced resolution corresponding to the defined gain factor V, and also containing the DC voltage parts contained in the input signal ES. 
     When, as a consequence of transgressions of the dynamic range by the edited signal AS, a switch must be undertaken to the digital data corresponding to the input signal ES resulting from the analog-to-digital conversion, slight discontinuities can occur. These discontinuities result from and offset voltage, caused by the amplifier 11 being superimposed on the edited signal AS. These discontinuities can be avoided in a third operating mode of the apparatus. For interrelated samples of the input signal ES and of the edited signal AS, the calculating unit 6 in this operating mode calculates the difference, with correct operational sign, between the digital data corresponding to the edited signal AS and the digital product of the input signal ES and the multiplication factor F. From the differences calculated for a selected number of successive samples, for example 256, the calculating unit 6 forms the average value, taking their operational signs into consideration. The calculating unit 6 adds this average value to the digital product of the defined factor F and the digital data corresponding to the input signal ES which were most recently taken into consideration in the calculation of the average value. The aforementioned discontinuities caused by the offset voltage of the amplifier 11 are thus avoided. The average value formation described above is not absolutely necessary, and may be eliminated. 
     Dependent on whether the apparatus operates according to the operating mode shown in one of FIGS. 2, 3 or 4, the difference which is formed will be one of a number of different possibilities. One entry in the two values whose difference is determined will be either the product of the digital data corresponding to the input signal and the defined multiplication factor F, or the product of the defined factor F and the digitally filtered input signal ES which has been subjected to the second transfer function. The other entry in the formation of the difference will be either the digital data corresponding to the edited signal AS, or the sums obtained by the above-described addition of the digital data obtained from the filtered input signal ES after the application of the third transfer function thereto. There is also the possibility of subtracting the average value (or the differences if this average value is not formed) in a corresponding manner from the digital data corresponding to the edited signal AS. 
     FIG. 5, which otherwise corresponds to FIG. 4, the above-described correction of the offset voltage of the amplifier 11 is symbolically illustrated by subtracting the amplified input signal ES from the output signal of the summing stage 33 at a further summing stage 34. The signal corresponding to the offset voltage of the amplifier 11 proceeds from the summing stage 34 to an RC element 36, having an integrating effect, formed by a resistor R1 and a capacitor C1 connected to ground. The average value is formed by this element 36. The average value stored in the capacitor C1 is added to the amplified input signal ES at a summing stage 35. As a consequence of the transgression of the dynamic range by the edited signal AS, the electronic switch 29 is actuated by the detector element 28 so that the signal proceeds from the summing stage 35 to the output A, with no discontinuity occurring. The two summing stages 34 and 35 and the RC element 36 may be present in an analogous way (not shown) in the embodiments of FIGS. 2 and 3 illustrating the other operating modes. 
     In a further operating mode of the apparatus, the signal generator 13, which generates a sawtooth signal PS, is used. In this operating mode, the electronic calculating unit 6 closes the switch 14 with an appropriate signal, so that the sawtooth signal PS is superimposed on the input signal ES at the summing stage 16. The maximum amplitude of the sawtooth signal PS is at least equal to the voltage difference which causes a change in the output data of the analog-to-digital converter 4 by one LSB. The duration of the period of the sawtooth signal PS is selected so that the analog-to-digital converter 4 undertakes a defined number of analog-to-digital conversions during the curve of the sawtooth signal PS along its linear edge S from one extreme value to the other extreme value of its amplitude. The defined number of analog-to-digital conversions is preferably equal to the defined gain factor V. In the described exemplary embodiment, it thus amounts to 32. 
     As a consequence of the superimposing of the sawtooth signal PS on the input signal ES in the summing stage 16, a step signal arises at the output of the sample and hold circuit 9, which proceeds via the multiplexers 1 and 3 to the analog-to-digital converter 4. The output data of the analog-to-digital converter 4 will change by at least one LSB during the course of the sawtooth signal PS along its linear edge S, leaving out of consideration the special case wherein the maximum amplitude of the signal PS is equal to the voltage difference which causes a change in the output data of the analog-to-digital converter 4 by one LSB, and also leaving out the special cases wherein the minimum or maximum amplitude of the signal obtained by the superimposition of the signal PS on the signal ES exactly corresponds to a voltage value measurable with the analog-to-digital converter 4. 
     The calculating unit 6 subjects the output data of the analog-to-digital converter 4, resulting from the analog-to-digital conversion of the input signal ES with the sawtooth signal PS superimposed thereon, to a fourth transfer function having an integrating effect. The integral of the output data of the analog-to-digital converter is thus formed for the output data which most recently continuously occurred in the defined plurality of analog-to-digital conversions. In this integration, the calculating unit 6 takes the maximum amplitude of the sawtooth signal PS, and the position of the sawtooth signal PS relative to the input signal ES after superimposition, into consideration as boundary conditions. 
     As a result of the integration, and thus as a result of the analog-to-digital conversion of the input signal ES, the calculating unit 6 then supplies digital data having a number of bits greater than the number of bits of the analog-to-digital converter. The number of bits thus increases corresponding to that number of analog-to-digital conversions which are undertaken during the time wherein the amplitude of the sawtooth signal PS along its linear edge changes by a value leading to a change in the output data of the analog-to-digital converter 4 by one LSB. If, as in the case of the described exemplary embodiment, the maximum amplitude of the sawtooth signal PS is equal to one LSB, and if 32 analog-to-digital conversions are undertaken during one period of the sawtooth signal PS, the number of bits of the results of the analog-to-digital conversion obtained according to this operating mode is increased from 14 to 19 bits. 
     The just-described operating mode of the apparatus is illustrated in FIG. 6, which is a diagram wherein a constant input signal ES is symmetrically superimposed with a sawtooth signal PS, entered with a dot-dash line, having an amplitude of 1.5 LSB. Six analog-to-digital conversions are undertaken during one period of the sawtooth signal PS. This means that four analog-to-digital conversions are undertaken during that time wherein the amplitude of the sawtooth signal PS changes by one LSB. As shown with reference to six typical integration fields, the integration respectively occurs over the output data D of six successive analog-to-digital conversions. In the integration, the calculating unit 6 first forms the sum H of six successively arising output data of the analog-to-digital converter 4. The sum H is identified by the area shaded by lines rising from left to right in FIG. 6. The number K of more significant output data of the analog-to-digital converter 4 is determined, on the basis of which the calculating unit 6 calculates the areas identified by shading with lines descending from left to right in FIG. 6. This takes place on the basis of an algorithm which is adapted to the particular conditions, which those skilled in the art can easily locate. For each integration event, the calculating unit 6 adds the values H and K, and divides them by the number of analog-to-digital conversions which were undertaken in the duration of the period. The result of the division corresponds to the result of the analog-to-digital conversion and, in the embodiment of FIG. 6, amounts to 16.25. 
     It can be seen that the signal which is superimposed on the input signal ES may be a signal of arbitrary wave form such as a noise signal, as long as it has a statistically uniform amplitude distribution. 
     In a last operating mode which serves the purpose of calibrating or balancing the apparatus, the sawtooth signal PS is supplied to the summing stage 17 by the calculating unit 6 closing the electronic switch 15, the electronic switch 14 being open. The sawtooth signal PS is thus superimposed both on the input signal ES and on the edited signal AS. Analog-to-digital conversion of the input signal ES ensues in the manner described above in conjunction with FIG. 6 having an increased number of bits, while the edited signal AS, is subjected to the third transfer function according to FIG. 4. The calculating unit 6 also undertakes the correction of the offset voltage of the amplifier 11 as set forth in conjunction with FIG. 5. An arbitrary signal, which need only satisfy the condition of not transgressing the dynamic range with respect to the edited signal AS, is then supplied to the apparatus. It is also possible to short-circuit the input of the apparatus, so that the input signal ES has the value of zero. The calculating unit 6 then compares the results of the analog-to-digital conversion, calculated by using the digital data corresponding to the edited signal AS, to the results of the analog-to-digital conversion calculated only using digital data corresponding to the input signal ES and multiplied by the factor F, without considering the digital data corresponding to the edited signal AS. A difference value is obtained as a result of this comparison. In the ideal case, the differences will reproduce the curve of the sawtooth signal PS, because this is interpreted as a single component when calculating the results of the analog-to-digital conversion using the digital data corresponding to the edited signal AS, but is only employed as an auxiliary signal in the case of the analog-to-digital conversion of the input signal ES. When the differences formed by the calculating unit 6 significantly deviate from the sawtooth signal PS, adjustment or balancing of the apparatus can be undertaken. Such adjustment or balancing may be achieved, for example, by adapting the third transfer function T3 to the component values of the high-pass filter 10, by correcting the gain factor V if this is variable, or by correcting the defined factor F. 
     Although modifications and changes may be suggested by those skilled in the art, it is the intention of the inventors to embody within the patent warranted hereon all changes and modifications as reasonably and properly come within the scope of their contribution to the art.