Abstract:
A transceiver coupled to an antenna includes: (a) a multi-port filter having a bidirectional port coupled to the antenna, at least one input port and at least one output port; (b) a transmit datapath receiving a transmission signal and providing the transmission signal for transmission by the antenna through the multi-port filter, the transmit datapath being coupled to the input port of the multi-port filter; (c) a receive datapath receiving a reception signal from the antenna, the receive datapath being coupled to the output port of the multi-port filter; (d) a band-pass filter coupled to the antenna for receiving a sampled signal that includes intermodulation components between two or more of an external signal, the transmission signal and the reception signal; and (e) a monitoring and cancellation circuit receiving the transmission signal, the reception signal and the sampled signal to cancel the intermodulation components.

Description:
BACKGROUND OF THE INVENTION 
       [0001]    1. Field of the Invention 
         [0002]    The present invention relates to a transmitter-receiver of a wireless communication system. In particular, the present invention relates to cancellation of passive intermodulation interferences in a transmitter-receiver of a wireless communication system. 
         [0003]    2. Discussion of the Related Art 
         [0004]    In a wireless base station, the sensitivity of an uplink receiver can be severely degraded by undesirable interfering signals within the uplink frequency band.  FIG. 1  is a block diagram of base-station station  100  in a wireless communication system using frequency-division-duplexed (FDD), with antenna  101  being shared between downlink transmitter  102  and uplink receiver  103 . As shown in  FIG. 1 , duplexer  104  is a three-port radio-frequency (RF) filter that isolates reverse-direction receiver signal  110  from forward-direction transmitter signal  111 . Forward-direction transmitter signal  111 , which is provided by power amplifier (PA)  105 , is typically a high-power RF signal. From duplexer  104 , transmitter signal  101  is sent into free space by antenna  101 . Nonlinear junctions in degraded antenna components (e.g. connectors) and rusty objects near antenna  101  (e.g. metal fences) introduce passive-intermodulation (PIM) interference in the reverse direction (i.e., back to receiver  102 ). The PIM effect is thus also known as “the rusty-bolt effect” and has been recognized by the wireless industry as a difficult and complicated problem. 
         [0005]    PIM interferences may be detected during antenna installation with currently available instruments using a high-power, two-tone signal. However, PIM problems often surface as a result of gradual degradation only years after installation. Carrying out instrument-based PIM test interrupts the service. Furthermore, when a base station has begun service, an instrument test may not be allowable because the PIM test sends a high-power RF signal into free space in a frequency band that is not licensed to the service provider. 
       SUMMARY 
       [0006]    According to one embodiment of the present invention, a transceiver coupled to an antenna includes: (a) a multi-port filter having a bidirectional port coupled to the antenna, at least one input port and at least one output port; (b) a transmit datapath receiving a transmission signal and providing the transmission signal for transmission by the antenna through the multi-port filter, the transmit datapath being coupled to the input port of the multi-port filter; (c) a receive datapath receiving a reception signal from the antenna, the receive datapath being coupled to the output port of the multi-port filter; (d) a band-pass filter coupled to the antenna for receiving a sampled signal that includes intermodulation components between two or more of an external signal, the transmission signal and the reception signal; and (e) a monitoring and cancellation circuit receiving the transmission signal, the reception signal and the sampled signal to cancel the intermodulation components. 
         [0007]    According to one embodiment of the present invention, the intermodulation components may be intermodulation between the transmission signal and the reception signal or between the external signal and the transmission signal. The external signal may have a frequency component with a predetermined frequency band of the transmission signal, or the external and the transmission signal may have different non-overlapping frequency bands. 
         [0008]    According to one embodiment of the present invention, the sampled signal may be obtained through a coupler tapping the bidirectional port of the multi-port filter or tapping the input port of the multi-port filter. Alternatively, the sampled signal is obtained from a second output port of the multi-port filter. 
         [0009]    According to one embodiment of the present invention, the transmit datapath may include a crest factor reduction processor. The transmit datapath may include: (i) a digital-to-RF converter that converts the transmission signal into an RF signal; (ii) an analog linearizer for providing pre-distorted RF signal; and (iii) a power amplifier for amplifying the pre-distorted RF signal for transmission by the antenna. Alternatively, the transmit datapath may include (i) a digital signal processor that receives the transmission signal to provide a pre-distorted transmission signal; (ii) a digital-to-RF converter that converts the pre-distorted transmission signal into an RF signal; and (iii) a power amplifier for amplifying the RF signal for transmission by the antenna. 
         [0010]    According to one embodiment of the present invention, the monitoring and cancellation circuit includes an intermodulation cancellation circuit which models non-linearity in the intermodulation components. In some embodiments, the non-linearity is modeled according to a polynomial function. In some embodiments, the non-linearity is modeled according to a rational approximation technique. The intermodulation cancellation circuit may include delay elements for aligning in time at least two of: the transmission signal, the reception signal and the sampled signal. The intermodulation cancellation circuit may include a low-pass filter and one or more local oscillators. In one embodiment, one of the local oscillators has a frequency corresponding the uplink-downlink frequency spacing. In another embodiment, the local oscillator has a frequency corresponding to the spacing between the transmission signal and the external signal. 
         [0011]    According to one embodiment of the present invention, the intermodulation cancellation circuit operates according to a set of adaptively adjusted parameters. The parameters of the intermodulation cancellation circuit are adjusted based on minimization of a cost function. The cost function may be minimized according to a mean-square-error criterion or a weighted mean-square of an error signal or its spectrum. Alternatively, the cost function may be minimized according to a quality-of-signal figure of merit. The quality-of-signal figure of merit corresponds to a signal-to-interference ratio or to an error vector magnitude. 
         [0012]    The present invention is better understood upon consideration of the detailed description below and the accompanying drawings. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0013]      FIG. 1  is a block diagram of base-station station  100  in a wireless communication system using frequency-division-duplexed (FDD), with antenna  101  being shared between downlink transmitter  102  and uplink receiver  103 . 
           [0014]      FIG. 2  is a block diagram of FDD-mode wireless transceiver system  200  that supports PIM monitoring and cancellation, in accordance with an embodiment of the present invention. 
           [0015]      FIG. 3  is a block diagram of PIM monitoring and cancellation subsystem  208 , in accordance with one embodiment of the present invention. 
           [0016]      FIG. 4  shows a generalized implementation of PIM canceller  302 , according to one embodiment of the present invention. 
           [0017]      FIG. 5  shows the variation of the 3 rd -order PIM power, under measurement and two analytical models, as a function of average input power for a SMC connector with balanced two-tone input. 
           [0018]      FIG. 6  is a schematic representation of digital circuit  600  for a memoryless nonlinearity model. 
           [0019]      FIG. 7  shows PIM canceller  700 , which is one implementation of PIM canceller  302 , according to one embodiment of the present invention. 
           [0020]      FIG. 8  shows circuit  800 , which is one implementation of Class-A PIM model  701  in PIM canceller  700  of  FIG. 7 , according to one embodiment of the present invention. 
           [0021]      FIG. 9  shows circuits  900  and  950 , which provide one implementation of Class-B/C PIM model  702  in PIM canceller  700  of  FIG. 7 , according to one embodiment of the present invention. 
           [0022]      FIG. 10  shows wireless transceiver system  1000 , which taps an external signal using a coupler placed between the PA and the terminal duplexer in a Class-B PIM model, according to one embodiment of the present invention. 
           [0023]      FIG. 11  shows using wireless transceiver system  1100  having a triplexer in a Class-C PIM model, as the local signal and the external signal are at different downlink bands, according to one embodiment of the present invention. 
           [0024]      FIG. 12  shows wireless transceiver system  1200  using digital pre-distortion techniques to linearize a power amplifier, in accordance with one embodiment of the present invention. 
       
    
    
       [0025]    Like elements in the figures are assigned like reference numerals. 
       DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       [0026]    In an FDD system, the inventors recognize that there are at least three classes of PIM interferences. The first class—“Class-A PIM interference”—is caused by self-mixing of a PA output signal at nonlinear junctions. In this detailed description, BW denotes the bandwidth of desired transceiver signals and f R  denotes the downlink/uplink frequency spacing. For example, in a base station that operates in band #2 under 3GPP Long Term Evolution (LTE), where the downlink frequency range is 1930˜1990 MHz, the uplink frequency range is 1850˜1910 MHz, f R =80 MHz and BW≦60 MHz. Thus, in this system, when the condition BW&gt;f R /(m+1) is satisfied, the (2m+1)-th order intermodulation falls into the receiver band. In such a system, the Class-A PIM is problematic for FDD bands that have relatively large bandwidth and relatively small downlink/uplink spacing. 
         [0027]    Among the different orders of PIM effects, the 3 rd -order effect has the highest intermodulation power. The following table shows the LTE bands (as defined in 3GPP TS36.104) where the 3 rd -order PIM can occur. 
         [0000]    
       
         
               
               
               
               
               
             
               
               
               
               
               
             
           
               
                 TABLE 1 
               
               
                   
               
               
                 LTE 
                 Downlink 
                 Uplink 
                 Maximum 
                 Frequency 
               
               
                 operating 
                 frequency 
                 frequency 
                 bandwidth 
                 spacing 
               
               
                 band # 
                 range (MHz) 
                 range (MHz) 
                 (MHz) 
                 (MHz) 
               
               
                   
               
             
             
               
                   
               
             
          
           
               
                 12 
                 729~746 
                 699~716 
                 17 
                 30 
               
               
                 5 
                 869~894 
                 824~849 
                 25 
                 45 
               
               
                 8 
                 925~960 
                 880~915 
                 35 
                 45 
               
               
                 3 
                 1805~1880 
                 1710~1785 
                 75 
                 95 
               
               
                 2 
                 1930~1990 
                 1850~1910 
                 60 
                 80 
               
               
                 7 
                 2620~2690 
                 2500~2570 
                 70 
                 120 
               
               
                   
               
             
          
         
       
     
         [0028]    PIM may be caused by the nonlinear interaction between a local PA output signal and an external signal from a nearby antenna, as a result of the external signal feeding into the local antenna by virtue of inter-antenna coupling. “Class-B PIM interference” occurs when the local PA output signal and the external signal occupy different portions of the same down-link frequency band. The LTE bands shown in Table 1 above are susceptible to class-B PIM interference. 
         [0029]    “Class-C PIM interference” occurs when the local PA output signal and the external signal belong to different downlink bands. Table 2 shows a list of FDD-mode LTE bands that can have Class-C PIM interference due to 3 rd -order PIM. For example, the 3 rd -order PIM between the LTE downlink band #4 (2110˜2155 MHz) and band #2 (1930˜1990 MHz) may fall into the uplink band #4 (1710˜1755 MHz). Reducing the bandwidth of the local PA signal may suppress class-A PIM interference, but would have little or no effect on Class-B and Class-C PIM interferences. 
         [0000]    
       
         
               
               
               
               
               
             
               
               
               
               
               
             
           
               
                 TABLE 2 
               
               
                   
               
               
                 LTE 
                 Local 
                 Local 
                 External 
                 External 
               
               
                 operating 
                 uplink 
                 downlink 
                 downlink 
                 downlink 
               
               
                 band # 
                 band (MHz) 
                 band (MHz) 
                 band #, mode 
                 range (MHz) 
               
               
                   
               
             
             
               
                   
               
             
          
           
               
                 1 
                 1920~1980 
                 2110~2170 
                 40, TDD  
                 2300~2400 
               
               
                 2 
                 1850~1910 
                 1930~1990 
                 34, TDD  
                 2010~2025 
               
               
                 3 
                 1710~1785 
                 1805~1880 
                 33, TDD  
                 1900~1920 
               
               
                 3 
                 1710~1785 
                 1805~1880 
                 2, FDD 
                 1930~1990 
               
               
                 4 
                 1710~1755 
                 2110~2155 
                 2, FDD 
                 1930~1990 
               
               
                 5 
                 824~849 
                 869~894 
                 8, FDD 
                 925~960 
               
               
                 6 
                 830~840 
                 875~885 
                 8, FDD 
                 925~960 
               
               
                 9 
                 1750~1785 
                 1845~1880 
                 2, FDD 
                 1930~1990 
               
               
                 10 
                 1710~1770 
                 2110~2170 
                 2, FDD 
                 1930~1990 
               
               
                 12 
                 699~716 
                 729~746 
                 14, FDD  
                 758~768 
               
               
                 14 
                 788~798 
                 758~768 
                 12, FDD  
                 729~746 
               
               
                   
               
             
          
         
       
     
         [0030]    The present invention mitigates PIM interferences by providing PIM monitoring and cancellation.  FIG. 2  is a block diagram of FDD-mode wireless transceiver system  200  that supports PIM monitoring and cancellation, in accordance with an embodiment of the present invention. As shown in  FIG. 2 , system  200  includes digital baseband integrated circuit  201  (“BBIC”), which performs baseband signal data processing. Mixed-signal transceiver integrated circuit  202  interfaces the digital processing domain and the RF or analog signal domain. Mixed signal transceiver integrated circuit  202  includes a transmitter portion with crest-factor reduction (CFR) processor  203 , which is followed by digital-to-RF up-converter  204 . Examples of crest factor reduction methods that are suitable for implementation in CFR processor  203  may be found in the copending U.S. patent application Ser. No. 13/897,119, entitled “Crest Factor Reduction for Band-Limited Multi-Carrier Signals” filed May 17, 2013. The disclosure of the &#39;719 application is hereby incorporated by reference in its entirety. Power amplifier (PA)  207  may be linearized by either predistortion techniques in the RF domain or digital predistortion techniques in baseband.  FIG. 2  shows RF power amplifier linearizer (RFPAL)  205 , which applies predistortion techniques to an RF signal to linearize power amplifier  207 . RFPAL  205  use techniques that are developed at Scintera Network, Inc., Santa Clara, Calif. As shown in  FIG. 2 , the linearized signal of PA  207  is provided to duplexer  210 , which is then transmitted into free space by antenna  211 . 
         [0031]    In  FIG. 2 , antenna  211  is also used to receive from free space an RF signal for mixed signal transceiver integrated circuit  202 . The RF signal to be received into the receiver portion of mixed signal transceiver integrated circuit  202  is forwarded by duplexer  210  to RF-to-digital down-converter  206 , after suitably amplified by low noise amplifier (LNA)  209 . RF-to-digital converter  206  provides as output a complex-valued digital signal denoted by r(t). The r(t) signal, referred to as the reception signal, contains the desired signal component, as well as superimposed PIM interferences. For example, Class-B or Class-C PIM interferences may be present as a result of an interaction between PA output signal  241  and an external signal. The external signal may feed into antenna  211  towards mixed signal transceiver integrated circuit  202 . In  FIG. 2 , coupler  212  placed between the output port of duplexer  204  and the input port of antenna  211  provides a copy of the antenna reverse signal. The coupling ratio of coupler  212  may be 20˜30 dB, for example. Band-pass filter (BPF)  213  suppresses undesired frequency components (e.g., antenna reflection in the frequency band of transmitter  102 ), but preserves the PIM interferences coupled from the external signal. Mixed signal transceiver integrated circuit  202  also includes PIM monitoring and cancellation subsystem  208 , which receives (i) transmit signal s(t) from CFR processor  203 , (ii) reception signal r(t) RF-to-digital down-converter  206 , (iii) the filtered reverse signal from the output terminal of BPF  213 , and (iv) control signal QoS (“quality of signal”) from digital baseband integrated circuit  201 . After canceling PIM interference, output signal y(t) is passed to digital baseband integrated circuit  201 . 
         [0032]      FIG. 3  is a block diagram of PIM monitoring and cancellation subsystem  208 , in accordance with one embodiment of the present invention. As shown in  FIG. 3 , PIM monitoring and cancellation subsystem  208  includes RF-to-digital down-converter  301 , which converts the filtered reverse signal of BPF  213  to a complex-valued digital signal denoted by x(t) (“down-converted reverse signal”). Thus, undesired frequency components in the antenna reverse signal is first removed by the RF-domain filtering in BPF  213 , followed by analog filtering, digital filtering or both in RF-to-digital down-converter  301 . PIM canceller  302  is a digital processor which receives transmit signal s(t), reception signal r(t), and down-converted reverse signal x(t), which are all complex-valued digital input signals. As shown in  FIG. 3 , the parameters of PIM canceller  302  are adaptively adjusted in parameter adaption circuit  303  based on the QoS signal to achieve best PIM cancellation performance under the current operating conditions. 
         [0033]      FIG. 4  shows a generalized implementation  400  for PIM canceller  302 , according to one embodiment of the present invention. In  FIG. 4 , τ s  delay block  401 , τ x  delay block  402  and τ r  delay block  403  are integer-sample delays, G i (·) block  408 - i  is i-th one of n memoryless nonlinearity models for the PIM sources, and δ i  delay block  404 - i , d i  delay block  405 - i , c i  block  406 - i  and summer  407 - i  form the i-th one of n interpolating FIR filters used to obtain fractional delays. c i  block  406 - i  is the i-th one of n complex gains. LPF block  412  is a low-pass filter that acts as a channel filter. Digital local oscillator  410  has frequency f R , which is the downlink-uplink frequency spacing (i.e., addressing intermodulation between the signal to be received and the signal to be transmitted). Digital local oscillator  415  has frequency f X , which is the external-transmit frequency spacing (i.e., addressing intermodulation between the signal being transmitted and the external signal). Therefore, output signal y(t) of PIM canceller  302  may be expressed as: 
         [0000]        y ( t )= r ( t−τ   r )+LPF{ e   j2πf     R     t   Σi   =1   2   G   i ( s ( t−τ   s −δ i )+ c   i   e   j2πf     X     t   x ( t−τ   x   −d   i ))}  (1)
 
         [0034]    Conventional odd-order polynomial models are often inadequate to provide good modeling of PIM source nonlinearity. According to the present invention, two modeling methods are provided for modeling PIM source nonlinearities. In the first method—the rational approximation model—the PIM source nonlinearity is expressed as: 
         [0000]    
       
         
           
             
               
                 
                   
                     
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         [0035]    where z denotes a complex-valued input to the PIM source nonlinearity model, a i  is a complex-valued parameter, ε i  and μ i  are non-negative real-valued parameters (typically, μ i /ε i &lt;0.05). Alternatively, in the second method—3 rd -and-4 th -order polynomial model—the PIM nonlinearity is expressed as: 
         [0000]        G   i ( z )= a   i   |z|   2   z+b   i   |z|   3   z   (3)
 
         [0000]    where a i  and b i  are complex-valued parameters. 
         [0036]      FIG. 5  shows the variation of the 3 rd -order PIM power, under measurement and two analytical models, as a function of average input power for a SMC connector with balanced two-tone input. The rational approximation model (model #2 in  FIG. 5 ) is surprisingly accurate even for such a worst-case connector that has a PIM power of −65 dBm (at 43 dBm input). The 3 rd -and-4 th -order polynomial model (model #1) is found applicable when the PIM power is lower than −85 dBm. 
         [0037]      FIG. 6  is a schematic representation of digital circuit  600  for a memoryless nonlinearity model. As shown in  FIG. 6 , envelope circuit  601  provides an envelope of the complex input value z, which is used to obtain a non-linearity value under the implemented model from look-up table (LUT)  603 . PIM nonlinearity G i (z) is the complex product computed in complex multiplier  602  between input value z and the model value from LUT  603 . LUT  603  is programmed to hold pre-computed values for the implemented non-linearity model. 
         [0038]      FIG. 7  shows PIM canceller  700 , which is one implementation of PIM canceller  302 , according to one embodiment of the present invention. In PIM canceller  700 , Class-A interference and Class-B or Class-C interferences are cancelled in Class-A PIM model  701  and Class-B/C PIM model  702 . 
         [0039]      FIG. 8  shows circuit  800 , which is one implementation of Class-A PIM model  701  of  FIG. 7 , according to one embodiment of the present invention. Circuit  800  is a particularized implementation of circuit  400  of  FIG. 4 , particularized for Class-A PIM interferences. Circuit  800  assumes that there are two PIM sources. Using the rational approximation nonlinearity model, the memoryless non-linearities of B 1 (·) and B 2  (·) provide: 
         [0000]    
       
         
           
             
               
                 
                   
                     
                       
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         [0040]    Alternatively, using the 3 rd -and-4 th -order polynomial model, the memoryless nonlinearity of B 1 (·) and B 2 (·) may be: 
         [0000]        B   1 ( x )=| x|   2   x, B   2 ( x )=| x|   3   x   (5)
 
         [0041]      FIG. 9  shows circuits  900  and  950 , which provide one implementation of Class-B/C PIM model  702  in PIM canceller  700  of  FIG. 7 , according to one embodiment of the present invention. In circuits  900  and  950 , only the 3 rd -order intermodulation between the transmit signal and the external signal is taken into account. In this embodiment, circuit  900  addresses PIM interference that occurs at f X ≈f R  and circuit  950  addresses PIM interference occurring at f X ≈f R /2. The inventors recognize that PIM interference occurs frequently at f X ≈f R  and at f X ≈f R /2. As shown in  FIG. 9(   a ), to isolate the interference at f X ≈f R , digital local oscillator  907  is set to frequency Δf=f R −f X . Similarly, to isolate the interference at f X ≈f R /2, digital local oscillator  957  is set to frequency Δf=f R −2f X . 
         [0042]    The PIM cancellers of the present invention may be adaptively controlled using monitoring and adjustable control parameters. The adaptive control technique adjusts the control parameters by optimizing a cost function, for example. One example of applicable adaptive control technique is disclosed, for example, in U.S. Pat. No. 8,136,081, entitled “Method and Apparatus to Optimize Adaptive Radio-Frequency Systems,” issued Mar. 13, 2012. The disclosure of the &#39;081 patent is hereby incorporated by reference in its entirety. In one embodiment, the cost function may be constructed using one of two methods, for example. 
         [0043]    The first method uses the PIM canceller&#39;s output signal, y(t), as an error signal for parameter identification and minimizes error signal y(t) under a mean-square-error (MSE) criterion. For example, the cost function may be the average power of the PIM canceller&#39;s output signal (i.e., conventional MSE), or a weighted mean-square of the error signal, ∫W(f)Y(f)df, with Y(f) being the power spectrum of error signal y(t) and W(f) a spectral window. 
         [0044]    Alternatively, under the second method, the cost function may be a quality-of-signal (QoS), e.g. the signal-to-interference ratio or the error vector magnitude (EVM) of reference symbols, obtained from demodulating the PIM canceller output signal in the BBIC. For example, in  FIG. 2 , baseband processor  201  provides a QoS signal based on PIM canceller  208 &#39;s output error signal y(t). 
         [0045]    In addition, in Class-A PIM model  702  ( FIG. 8 ) for PIM canceller  700  ( FIG. 7 ), the least-mean-square (LMS) algorithm provides an adaptation method for the complex-valued FIR coefficients. The LMS algorithm utilizes the correlation between the PIM canceller&#39;s output signal and a FIR tap output signal to adjust the corresponding FIR coefficients. In a Class-B PIM model (e.g., circuit  900  of  FIG. 9 ), where intermodulation occurs between the local transmit signal and the external signal within the same downlink band, the external signal may be tapped from a coupler placed between the PA and the terminal duplexer, as shown in wireless transceiver system  1000  of  FIG. 10 . 
         [0046]    In a Class-C PIM model, as the local signal and the external signal are at different downlink bands, a triplexer may be used to provide the reverse signals in two different bands.  FIG. 11  shows a system in which triplex  1101  include reception ports for extracting signals in the uplink receiver band and the external-signal band, respectively. 
         [0047]    The PIM monitoring and cancellation methods of the present invention are applicable in a transceiver system in which the PA is linearized by digital pre-distortion (DPD) techniques.  FIG. 12  shows wireless transceiver system  1200  using digital pre-distortion techniques to linearize a power amplifier, in accordance with one embodiment of the present invention. Applicable digital pre-distortion techniques for wireless transceiver system  1200  are disclosed, for example, in copending U.S. patent application Ser. No. 14/166,422, entitled “Adaptively Controlled Digital Pre-distortion in an RF Power Amplifier Using an Integrated Signal Analyzer with Enhanced Analog-to-Digital Conversion,” filed on Jan. 28, 2014. The disclosure of the &#39;422 patent application is hereby incorporated by reference in its entirety. Using the techniques disclosed in the &#39;422 patent application, the data processor in DPD subsystem  1202  is very low-power and occupies very little chip area. Hence, DPD subsystem  1202  may be easily integrated on mixed signal transceiver integrated circuit  1203 . The parameters of DPD subsystem  1202  may be controlled by signal analyzer  1201 . Signal analyzer  1201  need not be integrated onto mixed signal transceiver integrated circuit  1203 , as the analog-to-digital converters for PA monitoring have very different requirements from those used in uplink reception. 
         [0048]    The detailed description above is provided to illustrate the specific embodiments of the present invention and is not intended to be limiting. Numerous modifications and variations within the scope of the present invention are possible. The present invention is set forth in the accompanying claims.