Abstract:
A low phase noise frequency synthesizer includes arranged in series, a first mixer receiving a reference signal at a reference frequency F r , a loop filter and a voltage-controlled oscillator delivering a microwave signal at a second frequency F O  slaved to a multiple of reference frequency F r , the frequency synthesizer further includes: means of multiplication of the frequency F O  of the microwave signal by a factor N strictly greater than 1, means of correction of the frequency N·F O  of the output signal of the multiplication means to restore frequency N·F O  to an interval [F Omin , F Omax ] where output frequency F O  would vary if multiplication factor N=1, means of division of the frequency F j  of the output signal of the correction means by a factor equal to the expected ratio between frequency F j  and reference frequency F r , the frequency division means connected at output to the second input of the first mixer.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application is a National Stage of International patent application PCT/EP2012/054922, filed on Mar. 20, 2012, which claims priority to foreign French patent application No. FR 1100960, filed on Mar. 31, 2011, the disclosures of which are incorporated by reference in their entirety. 
     FIELD OF THE INVENTION 
     The present invention relates to the field of the generation of microwave signals by frequency-agile synthesizers of microwave signals, and more particularly to synthesizers based on the use of a phase-locked loop to slave the microwave signal at the desired frequency to a reference signal. 
     BACKGROUND 
     The generation of a microwave signal at a desired frequency is usually implemented using a circuit with a phase-locked loop. Such a circuit makes it possible to slave the frequency of the output signal to a multiple of the frequency of the reference signal. The output frequency may thus be chosen from several values solely by modifying the frequency division value of the feedback loop. One drawback of such a circuit is that it generates significant phase noise on the output signal. In fact, the output phase noise is increased by a factor equal to the division factor of the phase loop compared with the phase noise of the reference signal. 
     SUMMARY OF THE INVENTION 
     The invention provides an indirect frequency synthesizer, based on the principle of a phase-locked loop, but enabling a considerable decrease in the phase noise affecting the output signal while conserving the switching speed of the phase-locked loop between two frequency hops and while conserving the spectral purity of the signal generated. The invention also makes it possible to improve the frequency granularity of the signal generated. 
     The invention may be applied advantageously in all types of systems requiring the generation of frequency-agile microwave signals of high spectral purity and low phase noise. In particular the invention applies to the generation of radar transmission signals and to the frequency synthesizers used in metrology, as well as to the clock circuits of analog-to-digital or digital-to-analog converters. 
     The subject of the invention is thus a frequency synthesizer, with low phase noise, containing, arranged in series, a first mixer receiving at its first input a reference signal at a reference frequency F r , a loop filter and a voltage-controlled oscillator delivering at output a microwave signal at a second frequency F 0  and slaved to a multiple of said reference frequency F r , characterized in that it also comprises:
         means of multiplication of the frequency F 0  of said microwave signal by a factor N strictly greater than 1,   means of correction of the frequency N·F 0  of the output signal of said multiplication means configured to restore this frequency N·F 0  to an interval of variation [F 0min , F 0max ] in which the output frequency F 0  would vary if said multiplication factor N was equal to 1,   means of division of the frequency F j  of the output signal of said correction means by a factor equal to the expected ratio between said frequency F j  and the reference frequency F r ,   said frequency division means being connected at output to the second input of the first mixer.       

     In a particular aspect of the invention, the frequency correction means contain at least a second mixer, a plurality of local oscillators with low phase noise and a low-pass filter arranged in such a way that:
         the second mixer receives at a first input the output signal of said frequency multiplication means at a first frequency N·F 0 , and at a second input a signal delivered by one of said local oscillators of frequency F OLk  configured to correct said first frequency NF 0  to restore it to the interval of variation [F 0min , F 0max ] of the output frequency F 0 ,   said low-pass filter is configured to eliminate, in the output signal of said second mixer, the frequency components greater than the upper limit F 0max  of said interval [F 0min , F 0max ].       

     In a variant embodiment of the invention, the frequency value of the output microwave signal is obtained by the choice of one of the output signals of said local oscillators presented at the second input of said second mixer and by the choice of the frequency division value M j . 
     In a variant embodiment of the invention, said local oscillators are local dielectric resonator oscillators. 
     In a variant embodiment of the invention, the frequency F OLk  of the signals delivered by each local oscillator is determined, for k varying from 0 to N−1, by the following relationship: F OLk =(N−1)·F 0min +k·(F 0max −F 0min ) where k is equal to the integer part of the number 
               N   ·       (       M   i     -     M   1       )       (       M   2     -     M   1       )         ,         
with M 1  the ratio between the lowest output frequency F 0min  and the reference frequency F r , M 2  the ratio between the highest output frequency F 0max  and the reference frequency F r , and M i  the ratio between the desired output frequency F 0  and the reference frequency F r .
 
     In a variant embodiment of the invention, the division factor M j  of the means ( 208 ) is determined using the following relationship M j =N·M i −[(N−1)·M 1 +k·(M 2 −M 1 )]. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Other features and advantages of the invention will appear in the following description, made with reference to the appended drawings which represent: 
         FIG. 1 , a block diagram of a phase-locked loop according to the prior art, 
         FIG. 2 , a block diagram of the indirect frequency synthesizer device according to the invention, 
         FIG. 3 , a diagram illustrating the determination of the frequency correction value introduced into the feedback loop of the device according to the invention, 
         FIG. 4 , an illustration of the decrease in the phase noise on the microwave signal generated by comparing the performance of the known solutions and that of the invention. 
     
    
    
     DETAILED DESCRIPTION 
       FIG. 1  illustrates by a block diagram the principle of a phase-locked loop  100  enabling the slaving of a microwave signal S 0  of frequency F 0  to a multiple of the frequency F r  of a reference signal S r . 
     The reference signal S r  is compared by way of a mixer  101  to a signal S i  resulting from the frequency division  104  of the output microwave signal S 0  by a factor M i . The signal produced at the output of the mixer  101  contains the information on the error, of phase or of frequency, between the two signals it receives at input. This output signal is then filtered  102 , and supplied as input to a voltage-controlled local oscillator  103  that produces at its output the microwave signal S 0  of which the frequency is equal to the frequency F r  of the reference signal that the factor M i  multiplies. The assembly composed of the mixer  101  and of the filter  102  performs the function of comparator of the phase or the frequency between the reference signal S r  and the signal S i . The closed-loop operation guarantees a convergence of the system toward an output signal of which the frequency is such that the phase/frequency error at the mixer  101  output is approximately nil, or as close to nil as component defects will allow. 
     Thus, by varying the value M i  of the frequency division  104 , it is possible to choose the frequency of the output signal S 0  from an interval of values [F 0min , F 0max ]=[M 1 F r , M 2 F r ] 
     The phase noise, expressed in dBc/Hz, affecting the output signal S 0 , in the frequency band equal to the band of the loop filter  102 , is equal to θ O =θ ref +20·log(F 0 /F r )=θ ref +20·log(M i ) where θ ref  represents the sum of the phase noises of the reference signal and of the mixer  101 . The term phase noise is used here with reference to the power spectral density relative to the signal power. When the frequency F 0  is high with respect to the reference frequency F r , the integrated phase noise of the microwave signal S 0  becomes substantial. Generally speaking, the phase noise of the output signal is increased by the factor M i , compared to that of the reference frequency. The only way of reducing the phase noise then consists in increasing the frequency of the reference signal. However, such a modification is in most cases undesirable because it requires the changing of the reference signal generator circuit, most commonly a quartz oscillator, as well as of the frequency divider  104 . Moreover, the increase of the reference frequency also introduces the drawback of an increase in the granularity of the frequency resolution of the output signal. 
       FIG. 2  illustrates by a block diagram the indirect frequency synthesizer device  200  according to the invention. 
     The device  200  receives as input a reference signal S r  of frequency F r . The reference signal S r  is compared, by means of a mixer  201 , to a signal S i  output by the feedback path of the looped system  200  according to the invention. The signal resulting from the comparison of the signals S r  and S i  is then filtered by means of a loop filter  202  and presented as input to a voltage-controlled local oscillator  203 . The microwave signal S 0  at the desired frequency F 0 , a multiple of the frequency F r  of the reference signal, is obtained at the output of the local oscillator  203  that delivers a signal of frequency proportional to the voltage applied at its input. The assembly composed of the mixer  201  and of the filter  202  carries out the function of comparator of the phase or frequency between the reference signal S r  and the signal S i . 
     Part of the power of the microwave signal S 0  is then sampled and injected as input to a frequency multiplier  204  which produces as output a signal with frequency F 0  multiplied by a factor N. A mixer circuit  205  is connected at a first input to the output of the multiplier  204  and at a second input to a switch  206 , itself connected to one of N low phase noise local oscillators OL 1 , OL k , OL N . Each of said local oscillators OL k  delivers a signal with a frequency F OLk  predetermined as a function of the frequency F 0  of the microwave signal generated, as well as of the interval of variations [F 0min , F 0max ] of this frequency. Said local oscillators OL k  are, for example, DROs (Dielectric Resonator Oscillators) or PDROs (Phase locked Dielectric Resonator Oscillators). 
     The output signal of the mixer  205  contains at least one component at a frequency equal to the difference between the frequencies of the two signals applied to its input. This output signal is applied as input to a low-pass filter  207  with a cut-off frequency equal to F 0max  in order to retain only the useful frequency component and to filter the component corresponding to the sum of the frequencies of the two input signals. It is then frequency-divided by a factor M j  by a divider  208 , then applied to the second input of the mixer  201 . 
     One of the aims of the device  200  according to the invention consists in limiting the phase noise θ O  on the output signal S 0 , without modifying the frequency of the reference signal S r  or the values of the frequency divider  208  of the feedback loop. The introduction of the multiplier  204  makes it possible to reduce by a factor N the phase noise θ O , which is then equal to θ O =(M j /N) θ ref . 
     However, the introduction of the multiplier  204  changes the operation of a conventional phase-locked loop and it is appropriate to modify it to carry out the prime desired function, namely the synthesis of a microwave signal S 0  at a frequency F 0  that is a multiple, by a factor M i , of the reference frequency F r  and frequency-adjustable within an interval of variations [F 0min , F 0max ]=[M 1 ·F r , M 2 ·F r ]. In fact, the introduction of the multiplier  204  has the effect of increasing by a same factor the loop gain of the synthesizer. To correct this phenomenon, it is necessary to correct the frequency of the output signal of the multiplier  204 . The correction frequency F OLk  is determined in such a way as to restore the frequency of the output signal of the mixer  205  to the interval of variations [F 0min , F 0max ] expected by the divider  208  provided for conventional operation of the state of the art before the invention, i.e. when the multiplier  204  is absent or when the multiplication factor N is equal to 1. 
       FIG. 3  illustrates by a diagram an example of determining the correction frequency F OLk . On the frequency axis  300  is shown, firstly, the interval  301  of variation of the frequency of the microwave signal S 0  generated, and secondly the interval  302  of variation of the frequency of the output signal of the multiplier  204  itself decomposed into sub-intervals of identical width equal to the width F 0max −F 0min  of the interval  301 . The output signal of the frequency multiplier  204 , of frequency F i =N·M i ·F r , included in the interval [N·F 0min +k·(F 0max −F 0min ); N·F 0min +(k+1)·(F 0max −F 0min )], must be corrected by a frequency F OLk =(N−1)·F 0min +k·(F 0max −F 0min ) as illustrated by  FIG. 3 . 
     Thus, a signal of frequency F OLk  is generated for each value of k varying from 0 to N−1, by a separate local oscillator with low phase noise. 
     The output signal of the mixer  205  will by virtue of its construction have a frequency F j  lying within the interval  301  of variation of the output microwave signal S 0 . 
     The value of k is determined using the following relationship: 
               k   =     E   ⁡     [     N   ·       (       M   i     -     M   1       )       (       M   2     -     M   1       )         ]         ,       with   ⁢           ⁢     M   1       =           F     0   ⁢   min       /     F   r       ⁢           ⁢   and   ⁢           ⁢     M   2       =       F     0   ⁢   max       /       F   r     .                 
A frequency division  208  of a value M j =N·M i −[(N−1)·M 1 +k·(M 2 −M 1 )] is then applied to retrieve a signal, at the input of the phase comparator  201 , of frequency substantially identical to the reference frequency F r  in the steady state. In the transient state, the frequency F j  of the output signal of the mixer  205  tends gradually toward the product of the value of the division factor M j  and the reference frequency F r . In the steady state, this value becomes substantially equal to the ratio of the division factor value M j  and the reference frequency F r .
 
     The advantages of the invention are numerous when compared with known solutions. 
     First of all, the resultant phase noise on the generated microwave signal S 0  is reduced by a factor N with respect to a conventional phase-locked loop as described in  FIG. 1 . 
       FIG. 4  illustrates by two diagrams the phase noise generated on the output microwave signal of the device as a function of the bandwidth of the loop filter  102 ,  202 . 
     The left-hand part of  FIG. 4  represents the phase noise obtained for a conventional phase-locked loop. It is substantially equal to 20·log(M i ·θ r ) over the whole band of the loop filter  102 , M i  being the division factor of the loop and θ r  the phase noise of the reference signal. 
     The right-hand part of  FIG. 4  represents the phase noise obtained using the device according to the invention. It is decreased by a factor N over the whole frequency band under consideration except in a narrow band around the frequency F 0  of the generated signal, which corresponds to the loop band of the local oscillators OL k , typically of a width equal to a hundredth of the width of the loop band of the device. 
     The phase noise affecting the microwave signal is thus decreased in a very large part of the loop band of the device according to the invention. 
     The invention also has the advantage of not generating additional intermodulation spurs caused by the introduction of the second mixer  205 . In fact, these are filtered by the loop filter  202  and it is thus not necessary to implement a band-pass filtering at the output of the second mixer  205 ; a simple low-pass filter  207  is enough to eliminate the frequency component output from the mixer  205  which corresponds to the sum of the input frequencies. 
     The invention also has the additional advantage of improving the frequency resolution of the microwave signal generated. In fact, the step between two possible generated frequencies becomes equal to F r /N instead of F r  for a conventional phase-locked loop.