Abstract:
Provided are integrator circuit topologies that enable continuous integration without reset of the integrator circuit. One such integrator circuit includes a first integrator and a second integrator, each of the two integrators having a non-inverting terminal. Each of the non-inverting terminals is connected to an input node to alternately receive an input current for continuous integrator circuit integration without integrator circuit reset. The inverting terminal of the second integrator can be connected to an inverting terminal of the first integrator. The non-inverting terminal of the second integrator can be connected to an output of the first integrator through a first capacitor, and an output of the second integrator can be connected to a non-inverting terminal of the first integrator through a second capacitor. With such a capacitor connection, the capacitors alternately charge and discharge, based on integrator input current that is alternately directed between the non-inverting terminals of the integrators.

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
     This application is a divisional of copending U.S. application Ser. No. 09/502,134, filed Feb. 11, 2000, now issued as U.S. Pat. No. 6,380,790. 
    
    
     GOVERNMENT SUPPORT 
     This invention was made with Government support under Contract No. N65236-98-1-5407, awarded by DARPA. The Government has certain rights in the invention. 
    
    
     BACKGROUND OF THE INVENTION 
     This invention relates to integrators. 
     Integrators have high linearity, wide bandwidth, and low noise characteristics. Integrators, however, require a reset interval to discharge the capacitor in the integrator&#39;s feedback loop which results in significant “dead” times in measurements and harmful transients on the integrator&#39;s input. Additionally, the rapid discharge interval aggravates the problem of dielectric absorption, thereby undermining the lower limit of instrument precision. 
     Referring to FIG. 1, an integrator  10  includes a feedback loop having a switch  12  in parallel with a feedback capacitor  14 . The switch  12  allows the feedback capacitor  14  to discharge when the switch  12  is closed. Placing one or more strings of series resistors and capacitors in parallel with the feedback capacitor  14  with or without the switch  12  reduces at least some of the harmful effects of this discharge. However, even in some arrangements having multiple capacitors, dielectric absorption is still a problem since the charge in the series capacitors is redistributed with the feedback capacitor  14 . 
     SUMMARY OF THE INVENTION 
     The invention overcomes these unwanted effects of integrator reset by providing integrator circuit topologies that enable continuous integration, without the need for reset of the integrator circuit. One such integrator circuit includes a first integrator and a second integrator, each of the two integrators having a non-inverting terminal. Each of the non-inverting terminals is connected to an input node to alternately receive an input current for continuous integrator circuit integration without integrator circuit reset. 
     In further configurations, the inverting terminal of the second integrator can be connected to an inverting terminal of the first integrator. The non-inverting terminal of the second integrator can be connected to an output of the first integrator through a first capacitor, and the output of the second integrator can be connected to the non-inverting terminal of the first integrator through a second capacitor. In operation, the first integrator and the second integrator have voltages on respective ones of the inverting and non-inverting terminals that are substantially equal, and the two integrators produce output voltages that are complementary. 
     In a further integrator circuit provided by the invention, at least one integrator is provided, having an input for receiving an input current. A plurality of integrator feedback capacitors are provided, with each capacitor being connected to alternately charge and discharge, based on the integrator input current. This cooperative charging and discharging enables continuous integrator circuit integration without integrator circuit reset. 
     These integrator circuit topologies can be employed in a wide variety of applications in which low signal level, precise measurements are required. For example, in one biological application, the first integrator and the second integrator can be operated to each introduce an output voltage into a chemical bath on either side of a biological membrane. In this application, the integrator circuit is configured to detect fluctuations of ion channels. In another application, e.g., the integrator circuit can be configured for charge detection. 
     These applications are particularly well-served by the integrator circuit of the invention in its elimination of a need for rapid discharging of feedback capacitors during operation. The integrator circuit of the invention can perpetually integrate incoming current signals, such as low-level transducer signals, to produce an output of a continuous flow of two complementary voltages. This perpetual integration eliminates “dead time” and input transients, compensates for charge injection at the integrator input, and reduces the harmful effects of dielectric absorption. At the same time, the integrator circuit maintains a high degree of operational linearity, produces a low level of noise, and can accommodate a wide bandwidth of input signals. 
     Other features and advantages of the invention are provided in the following detailed description and the accompanying drawings, and in the claims. 
    
    
     DESCRIPTION OF DRAWINGS 
     FIG. 1 is a schematic diagram of a conventional integrator. 
     FIG. 2 is a block diagram of a chopper stabilizing circuit. 
     FIG. 3 is a schematic diagram of the block diagram of FIG.  2 . 
     FIG. 4 is a graph showing the output of the integrator circuit of FIG.  3 . 
     FIGS. 5-6 are graphs showing the chopper stabilization of the integrator circuit of FIG.  3 . 
     FIG. 7 is an unfolded view of the integrator circuit of FIG.  3 . 
     FIGS. 8-9 are graphs showing the output of the integrator circuit of FIG.  3 . 
     FIGS. 10-11 are graphs showing the response of the integrator of FIG. 3 to input current. 
     FIGS. 12-13 are graphs showing the charge injection compensation of the integrator circuit of FIG.  3 . 
     FIG. 14 is a graph showing currents detectable by the integrator circuit of FIG.  3 . 
     FIG. 15 is a graph showing charge detection by a conventional integrator circuit. 
     FIG. 16 is a graph showing charge detection by the integrator circuit of FIG.  3 . 
     FIG. 17 is a schematic diagram of the differentiator circuit of FIG.  2 . 
    
    
     DETAILED DESCRIPTION 
     Referring to FIG. 2, a chopper stabilizing circuit  20  includes a switching circuit  22 , an integrator circuit  24 , a sensing circuit  26 , a control circuit  28 , and a differentiator circuit  30 . In general, the chopper stabilizing circuit  20  has a topology and is controlled in a manner that eliminates the need for rapid discharging of feedback capacitors in the integrator circuit  24 . In particular, and as will be discussed in greater detail below, this advantage is accomplished by alternating the signal current from the switching circuit  22  to the integrator circuit  24 . In this way, the integrator circuit  24  can perpetually integrate these incoming current signals (low-level transducer signals) and output a continuous flow of two complementary voltages. The sensing circuit  26  detects when one of the complementary voltages reaches a threshold value and notifies the control circuit  28 . The control circuit  28  responds by sending a signal to the switching circuit  22 . This signal changes the position of switches in the switching circuit  22 , thereby alternating the signal current to the integrator circuit  24 . The differentiator circuit  30  receives the complementary voltages output by the integrator circuit  24  and provides a demodulated differentiation bit stream representing the slope of the complementary voltages. As will be described in more detail below, this chopper stabilizing circuit  20  eliminates dead time and input transients, compensates for charge injection at the input, and reduces the harmful effects of dielectric absorption. At the same time, the chopper stabilizing circuit  20  maintains high linearity, low noise, and wide bandwidth. 
     In the layout of the chopper stabilizing circuit  20 , the switching circuit  22  has an input at a first node  32  for receiving an input signal. The input signal includes the driving current/voltage for the chopper stabilizing circuit  20  from a load, a current source, and/or a voltage source. The switching circuit  22  has an output at a second node  34  that is determined by the position of the switch(es) included in the switching circuit  22 . The integrator circuit  24  has an input at the second node  34  for receiving an input signal from the switching circuit  22  and an output at a third node  36 . The sensing circuit  26  has an input at the third node  36  for receiving an input signal from the integrator circuit  24  and an output at a fourth node  38 . The control circuit  28  has an input at the fourth node  38  for receiving an input signal from the sensing circuit  26  and output at a fifth node  40  and a sixth node  42 . The switching circuit  22  has an input for receiving an input signal from the control circuit  28  at the fifth node  40 . This input signal controls the position of the switch(es) in the switching circuit  22 . 
     The differentiator  30  is shown in FIG. 2, though its presence is not necessary to ensure proper functioning of the chopper stabilizing circuit  20 . If it is not present, the integrator circuit  24  and the control circuit  28  may not necessarily have outputs at the third node  36  and the sixth node  42 , respectively. The differentiator circuit  30  has an input at the third node  36  for receiving an input signal from the integrator circuit  24  and at the sixth node  42  for receiving an input signal from the control circuit  28 . The input signal at the sixth node  42  controls the switch(es) included in the differentiator circuit  30 . The differentiator also has an output at a seventh node  44 . 
     Referring to FIG. 3, one particular embodiment of a chopper stabilizing circuit  20  includes a switching circuit  22 , an integrator circuit  24 , a sensing circuit  26 , and a control circuit  28 . The chopper stabilizing circuit  20  eliminates the need for rapid discharging of feedback capacitors  60   a-b  (preferably Teflon®) in the integrator circuit  24  by alternating the signal current from the switching circuit  22  to two integrators  62   a-b  included in the integrator circuit  24 . In this way, one feedback capacitor discharges while the other charges, thereby providing two inversely related output voltages (Vout+, Vout−) at Vout nodes  36   a-b . Once either of the output voltages reaches a predetermined threshold value (Vth), a regenerative comparator  76   a-b  included in the sensing circuit  26  and connected to this output voltage is tripped. Hysteresis prevents the sensing circuit  26  from causing false resets. The comparator  76   a-b  triggers a D-type flip-flop  78  through a NAND gate  79 , both included in the control circuit  28 . As the flip-flop  78  changes state, the outputs Q and Q-bar connected to the switches  66   a-b ,  68   a-b  cause them to reverse position. This reversal preserves the same orientation with respect to the load  72 , maintaining a uniform bias, while alternating the signal current to the integrator circuit  24 . 
     More specifically, the switching circuit  22  includes two pairs of two symmetric switches  66   a-b ,  68   a-b . The switches  66   a-b ,  68   a-b  may be any type of standard MOS (metal oxide semiconductor) switch, e.g., MAXIM 326. Only one set of switches  66   a-b ,  68   a-b  is closed at a time, each closed switch providing a path for a signal to the non-inverting input terminal of an operational amplifier (opamp)  70   a-b , e.g., Burr-Brown OP627, included in the integrators  62   a-b . When the phase one (&gt;1) switches  66   a-b  are closed, a load  72  provides the input current (Io) to the first opamp  70   a  while a voltage source  74  provides the bias voltage (Vb) to the second opamp  70   b . When the phase two (&gt;2) switches are closed, the load  72  and the voltage source  74  provide current/voltage to the other opamp  70   a-b . The values of Vout+ at the Vout node  36   a  and Vout− at the Vout node  36   b  depend on the position of these switches  66   a-b ,  68   a-b.    
     FIG. 4 shows the inverse relationship between Vout+ (Vcf2) and Vout− (Vcf1). In this scenario, the &gt;2 switches  68   a-b  begin closed and the feedback capacitors  60   a-b  initially are discharged, so Vout+ and Vout− begin at Vb. When Io flows through the load  72 , Vout+ and Vout− alternately and inversely ramp up and down in accordance with:               V          t       =       Io   Cf     .                            
     When Io decreases at a time t 1 , this relationship ceases. 
     The integrator circuit  24  can effectively integrate forever (constantly flowing Io), with negligible glitching during phase switching. This lack of glitch is helped by the symmetry of input stage of the integrator circuit  24 . Every input stage node  80   a-c  sees one switch  66   a-b ,  68   a-b  turn on and another turn off during a phase transition. The already low charge injection of the switches  66   a-b ,  68   a-b  is then effectively reduced to tens of femtoCoulombs (fC). Additionally, the symmetric pair requires no voltage drop across a switch  66   a-b ,  68   a-b , aiding in keeping leakage currents below a picoAmp (pA). The voltages at the input stage nodes  80   a-c  are substantially the same. 
     Referring to FIGS. 5 and 6, it is appreciated that offset may be a problem as in FIG. 5, but techniques exist to alleviate this problem, e.g., a stabilizing circuit. FIG. 5 shows the chop before stabilization, and FIG. 6 shows the chopper stabilization of the integrator circuit  24 . 
     Referring to FIG. 7, an unfolded view of the integrator circuit  24  helps demonstrate the manner in which the circuit functions. The compensation of the integrator circuit  24  may be broken down into two sections: minor and major loops. The minor loop concerns the stability of each opamp  70   a-b ; the major loop comprises the total feedback loop around the integrator. The major loop encompasses a unity gain inverter with a voltage divider formed by the first feedback capacitor  60   a  reacting with the capacitance off the input stage of the first opamp  70   a . The input capacitance is dominated by the opamp input capacitance and the parasitics of the switches  66   a-b ,  68   a-b . The ratio of the capacitive voltage divider in this embodiment is approximately ten, which will keep the major loop crossover well below that of the minor loops. The minor loops are stabilized with the addition of shunt capacitances  82   a-b , which help compensate for phase lag due to shunt resistors  84   a-b  (preferably metal film) reacting with the input capacitance of the opamps  70   a-b . With the bandwidth of the opamps  70   a-b  on the order of 10 MHz in this embodiment, the chopper stabilizing circuit  20  should be able to track currents with a bandwidth of approximately 1 MHz. 
     FIGS. 8-13 further demonstrate the functioning of the integrator circuit  24 . FIG. 8 shows Vout+ and Vout− with 50 μs per horizontal division, the typical reset duration in standard integrators, e.g., Axopatch  200 B and nuclear physics instrumentation. FIG. 9 shows a zoom in on the reset transient, with the switching occurring of the order of 500 ns, e.g., 700 ns. The 2 pF feedback capacitor  60   a-b  and a residual voltage jump of 20 mV signifies under 40 fC of charge injection. FIG. 10 shows the response of the integrator circuit  24  (top trace) to input current (bottom trace), a 2 nA peak-to-peak triangle wave. Because of this response, the integrator circuit  24  could be used for direct digitization of input current via single-slope integration by measuring the period between resets. FIG. 11 shows the response of the integrator circuit  24  in FIG. 10 superimposed with a 100 kHz sinusoid supplied by a 2 pF capacitor at the input. FIG. 12 shows the charge injection before compensation, and FIG. 13 shows the charge injection after compensation by the integrator circuit  24 . 
     Now referring to FIG. 14, the integrator circuit  24  can be used to detect the fluctuations of ion channels important in cell signaling and biological transport. These currents range from 0.1 pA to 100 pA, with bandwidths of 10 kHz. The integrator circuit  24  allows for measuring these currents without glitches from resetting. 
     Now referring to FIGS. 15 and 16, the integrator circuit  24  can also be used for charge detection. For example, x-ray and particle detectors output charge pulses that are usually integrated. Whenever a conventional integrator hits a limit value as in FIG. 15, it must reset and data can be lost. Using the integrator circuit  24 , the dead-time (lost data) is greatly reduced by the absence of capacitor resets as shown in FIG.  16 . 
     A differentiator circuit  30 , shown in FIG. 17, may be part of a chopper stabilizing circuit. The differentiator circuit  30  includes two switches  92   a-b . The switches  92   a-b  may be any type of standard MOS (metal oxide semiconductor) switch, e.g., MAXIM 326. Each switch  92   a-b  is either in a horizontal (&gt;1) position, e.g., switch  92   a  from a top start node  94   a  to a top end node  96   a , or a diagonal (&gt;2) position, e.g., switch  92   a  from the top start node  94   a  to a bottom end node  96   b , at any given time. Each closed switch  92   a-b  provides a path for a signal at entering nodes  36   a-b  to travel to the inverting terminal or to the non-inverting terminal of an opamp  100 . Input from a control circuit (not shown) determines the position of the switches  92   a-b . If the differentiator circuit is connected to the chopper stabilizing circuit  20  (see FIG.  2 ), the output from the control circuit  78  provides the phase information for the switches  92   a-b.    
     A number of embodiments of the invention have been described. Nevertheless, it will be understood that various modifications may be made without departing from the spirit and scope of the invention. Accordingly, other embodiments are within the scope of the following claims.