Abstract:
A compensation circuit (KS) is provided between the output of a differential amplifier (Diff_Amp) and the input of a controller (R). The compensation circuit generates a compensation signal, whose characteristic curve approximates to that of the parasitic signal, with the same amplitude and frequency (Phi1) as the parasitic signal, but by 180 DEG out of phase. The compensation signal is subtracted from the differential signal (DELTA Vs) and allows the parasitic signal to be eliminated to a great extent.

Description:
CROSS REFERENCE TO RELATED APPLICATION  
         [0001]    This application is a continuation of copending International Application No. PCT/DE01/02571 filed Jul. 12, 2000, which designates the United States.  
         BACKGROUND OF THE INVENTION  
         [0002]    The invention relates to a circuit arrangement for compensating interference signals in the control loop of a linear lambda probe according to the features of the preamble of claim 1.  
           [0003]    Legislators are using tax measures to promote the development of motor vehicles with lower fuel emissions and lower consumption of fuel.  
           [0004]    In spark ignition engines with stoichiometric mixture formation (λ=1) this has led to the development of SULEV vehicles (Super Ultra Low Emission Vehicles) with extremely low emissions.  
           [0005]    In order to save fuel, engines with direct high pressure injection of gasoline (HPDI, High Pressure Direct Injection) are currently being developed and introduced into the market. The fuel here is injected directly into the combustion space at increased pressure (approximately 150 bar). The conditioning of the mixture which is possible in this way can vary between rich, stoichiometric and lean. For the partial load operating mode of the engine a lean mixture formation offers considerable advantages in terms of consumption.  
           [0006]    Both developments require a significantly more precise control of the mixture than is possible with currently customary lambda probes (binary step change probes). In addition, by binary step change probes have an extremely restricted measuring range around λ=1. They are therefore unsuitable for measurements in the lean operating mode λ&gt;1.  
           [0007]    For this reason, lambda probes with an extended linear measuring range, which are referred to as linear lambda probes, and circuit arrangements for operating them are being increasingly used.  
           [0008]    [0008]FIG. 1 shows a linear lambda probe which is known per se. It has a heating element H, two electrode pairs VsC and IpC and a measuring chamber Mk which is connected to the exhaust gas stream A via a diffusion barrier GDP. The first electrode pair VsC is arranged between the measuring chamber Mk and air L and is used—similarly to the step jump probe—to measure the oxygen concentration in the measuring chamber Mk. The second electrode pair IpC is arranged between the measuring chamber Mk and the exhaust gas stream A. It permits—when a current Ip of appropriate polarity is applied—oxygen ions to be pumped out of the measuring chamber Mk or into it; hence the designation pump electrodes.  
           [0009]    It is thus possible to generate a dynamic equilibrium between the flow of oxygen through the diffusion barrier and the flow of oxygen ions through the pair of pump electrodes. A suitable controlling criterion here is the oxygen concentration in the measuring chamber Mk which is determined using the measuring electrodes. A preferred value is, for example, Vs=450 mV for λ=1.  
           [0010]    The pumping current Ip which flows in this case is a measure of the oxygen concentration in the exhaust gas. (And also of λ after numerical conversion).  
           [0011]    Some lambda probes require an artificial oxygen reference for operation. This is produced by pumping oxygen out of the measuring chambers to the positive reference electrode Vs+ by means of a small current Icp (for example 25 μA). The oxygen concentration which is produced as a result is then used for its part as a reference point for measuring the oxygen concentration in the measuring chamber Mk. The evaluation circuit must make this current available.  
           [0012]    The relationship between the oxygen concentration in the exhaust gas and the pumping current Ip is influenced by a number of probe parameters. For reasons of fabrication, the dynamic resistance of the diffusion barrier fluctuates. This would result in a deviation of the transmission ratio (gain errors). During fabrication, this is compensated by measuring and inserting a calibrating resistor Rc into the probe plug.  
           [0013]    [0013]FIG. 2 shows a basic circuit diagram of a known device for operating a linear lambda probe of an internal combustion engine.  
           [0014]    A first terminal Vs+, a second terminal Vp−/Vs−, a third terminal Vp+ and a fourth terminal Rc extend out of the probe S and are connected to the evaluation circuit. The probe heater and its terminals are not illustrated.  
           [0015]    The inverting input of a controller is connected to the first terminal Vs+ of the probe S and its noninverting input is connected to a center voltage Vm (Vm≈Vcc/2) via a reference voltage Vref, Vcc (usually 5 V) being a supply voltage of the circuit.  
           [0016]    The second probe terminal Vp−/Vs− and the inverting input of a pumping current source Ip Pump, whose noninverting input is connected to the output of the controller, are also connected to the center voltage Vm.  
           [0017]    The output of the pumping current source Ip Pump is connected to the fourth input Rc of the probe S.  
           [0018]    As the resistor Rc is subjected to considerable environmental loading owing to its installation position in the probe plug, a further resistor Rp is connected in parallel with it to the terminals Vp+ and Rc in the controller. This reduces the influence of a tolerance fault of Rc on the measuring accuracy of the pumping current Ip.  
           [0019]    The method of operation of the known circuit arrangement illustrated in FIG. 2 for operating a linear lambda probe (without generating Icp) is as follows:  
           [0020]    The terminal Vp−/Vs− of the probe is, like the reference voltage Vref, connected to the center voltage Vm. This serves as a reference voltage of the circuit.  
           [0021]    The control amplifier R compares the Nerst voltage Vs of the probe with the reference voltage Vref (for example 450 mV) and generates an output voltage which is converted by the subsequent pumping current source I Pump into a corresponding current Ip which then flows through the pumping cell to the center voltage Vm. The pumping current brings about a change in the oxygen concentration in the measuring chamber of the probe, which in turns results in a change in the Nernst voltage Vs. The difference between Vs and Vref (=ΔVs) constitutes the control error of the loop. The pumping current Ip can be measured as a voltage drop at the resistor Rp/Rc. It is used as measure of the oxygen concentration in the exhaust gas.  
           [0022]    In the stable control state (λ=1 in the measuring chamber), the Nernst voltage Vs is, for example, precisely 450 mV (ΔVs=0).  
           [0023]    Equilibrium prevails between the oxygen flow through the diffusion barrier and the oxygen ion flow, caused by the pumping current Ip. The maximum range of the output voltage of the pumping current I Pump ranges from approximately 0.1 V to 4.9 V.  
           [0024]    Alternatively, the control amplifier can also be embodied as an OTA (Operational Transconductance Amplifier) whose output stages form a current source. The output signal here is already a current and not—as is customary in the case of the operational amplifier—a voltage.  
           [0025]    Furthermore, the dynamic resistance of the diffusion barrier has a temperature dependence and pressure dependence illustrated in FIG. 3, which in turn causes faults in the transmission ratio. The temperature dependence is counteracted by measuring the probe temperature and controlling it by means of a heating element installed in the probe. For reasons of cost, a separate thermal element is not used here. Instead, the highly temperature-dependent internal resistance of the probe (probe impedance) is measured. The pressure dependence cannot be sensed in the probe by measuring equipment. If a separate pressure sensor is not used, an attempt is made to determine the dependence by means of a model-based calculation in the microcontroller and to compensate it numerically.  
           [0026]    [0026]FIG. 4 shows the probe impedance Ris of the probe S and its temperature dependence. The probe impedance can be represented as a temperature-dependent, complex impedance with a plurality of RC elements, in which case:  
           [0027]    R 1 /C 1  represents the contact resistance between electrodes and ceramic material,  
           [0028]    R 2 /C 2  represents the junction between the grain boundaries of the ceramic sintered elements, and  
           [0029]    R 3  represents the intrinsic resistance of the sintered material.  
           [0030]    As one of the electrodes of the pumping cell is subjected to the exhaust gas, its internal resistance changes to a very great extent. For this reason, the Nernst cell Vs is used to measure the probe temperature. Here too, R 1  changes and should therefore not be used for measuring temperature. As the time constant R 1 *C 1  has the highest value (lowest frequency), it is possible to reduce its influence by suitable selection of the measuring frequency. The impedance of the series connection of R 2 /C 2  and R 1  is therefore measured. The impedance Ris of a typical linear lambda probe is approximately 100Ω at a temperature of approximately 500° C. to 700° C. (and a measuring frequency of 3 kHz).  
           [0031]    Measurement of the internal resistance Ris:  
           [0032]    A known measuring method for determining Ris is to apply an alternating current, for example 500 μA (peak-to-peak, abbreviated below to ss) to the probe terminal Vs+. As a result of this alternating current, an alternating voltage of 500 μA*100Ω=50 mV (ss) is produced at Ris and it is superimposed on the Nernst voltage Vs, the actual probe signal, as an interference signal.  
           [0033]    [0033]FIG. 6 shows a typical voltage profile of the alternating voltage signal which forms an interference signal for the Nernst voltage. The signal is amplified in an amplifier V, for example by a factor 10, and then rectified in a rectifier GLR. The DC voltage Vri which has been produced in this way can then be fed to a microprocessor in order to control the temperature.  
           [0034]    For example, the alternating current is generated, as illustrated in FIG. 5, by means of a 3 kHz square-wave oscillator OSZ which is supplied with a voltage Vcc. The signal is conducted to the probe terminal Vs+ via a high-impedance resistor R 1  and a decoupling capacitor C1.  
           [0035]    A basic problem of this circuit arrangement is the abovementioned mutual influence between Vs and this interference signal as this interference signal also appears at the input of the controller and constitutes a control error. The controller will attempt to compensate this control error within the scope of its bandwidth. To do this, it changes the pumping current Ip, which in turn has effects on the Nernst voltage Vs. As the pumping current Ip is the measured variable for λ, the primary probe signal is falsified. FIG. 7 makes this fact apparent. In the case of the 3 kHz signal (upper signal), the peak and the pulse tilt are falsified by the Vs signal, and in the case of the Vs Signal (lower signal) the 3 kHz triangular signal is undesired.  
         SUMMARY OF THE INVENTION  
         [0036]    The object of the invention is to reduce the interference signal which is contained in the fault signal ΔVs and is undesired for the lambda control, in such a way that it no longer influences the pumping current control.  
           [0037]    This object is achieved according to the invention in that a compensation circuit is inserted into the pumping current circuit (FIG. 5).  
           [0038]    An embodiment of the present invention is a circuit arrangement for compensating interference signals in the control loop of a linear lambda probe of an internal combustion engine which is connected to an evaluation circuit which has a differential amplifier which forms the difference between the Nernst voltage which is measured in the lambda probe and subjected to an interference signal caused by the measurement of the probe impedance and a reference voltage which is related to a center voltage, and having a controller which generates a controlling voltage which is assigned to the difference and which is converted into a pumping current by a subsequent pumping current source, wherein a compensation circuit which generates a compensation signal which is approximated in its curve profile to the interference signal and has an amplitude and frequency which are identical to the interference signal but phase-shifted through 180°, which compensation signal is subtracted from the differential signal and as a result largely cancels out the interference signal, is provided between the output of the differential amplifier and the input of the controller.  
           [0039]    Another embodiment is a circuit arrangement for compensating interference signals in the control loop of a linear lambda probe comprising:  
           [0040]    an evaluation circuit with a differential amplifier which forms the difference between a Nernst voltage which is subjected to an interference signal and a reference voltage,  
           [0041]    a controller generating a controlling voltage from the difference wherein the controlling voltage is converted into a pumping current by a subsequent pumping current source, and  
           [0042]    a compensation circuit which generates a compensation signal which is approximated in its curve profile to the interference signal and has an amplitude and frequency which are identical to the interference signal but phase-shifted through 180°, wherein the compensation signal is subtracted from the differential signal and as a result largely cancels out the interference signal, coupled between the output of the differential amplifier and the input of the controller.  
           [0043]    The compensation circuit may contain an amplifier which is connected as a buffer and an integrator, the differential signal and the output signal of the integrator can be fed to the input of the buffer amplifier, and the output signal of the buffer amplifier can be fed to the inverting input of the integrator and to the input of the controller. The controller can be combined with the compensation circuit which contains an amplifier which is connected as an inverter and an integrator, the controller can be connected as a summing amplifier to two inputs, the differential signal can be fed to the one input of the controller, and the output signal of the controller can be fed to the input of the integrator whose output signal is fed to the second input of the controller via the inverter. The integrator may contain an integration capacitor whose polarity is reversed in synchronism with the oscillator frequency by means of an alternating switch, wherein, in the one—positive—phase, the one terminal of the integration capacitor is connected to the output of the integrator and the other terminal is connected to the inverting input of the integrator, and in the other—negative—phase the one terminal is connected to the inverting input and the other terminal is connected to the output of the integrator. The integrator may also contain at least two integration capacitors whose polarity is reversed in synchronism with the oscillator frequency by means of alternating switches in such a way that, in identical numbers of phase sections, corresponding to the number of integration capacitors, of the one—positive—phase, the first terminals of the integration capacitors are successively connected to the output of the integrator and other terminals are connected to the inverting input of the integrator, and in identical numbers of phase sections, corresponding to the number of integration capacitors, of the other—negative—phase, the first terminals are successively connected to the inverting input and the other terminals to the output of the integrator. The interference signal and the oscillator signal can be generated by means of the same signal source.  
           [0044]    A method for compensating interference signals in the control loop of a linear lambda probe may comprise the steps of:  
           [0045]    forming the difference between a Nernst voltage which is subjected to an interference signal and a reference voltage,  
           [0046]    generating a controlling voltage from the difference,  
           [0047]    converting the controlling voltage into a pumping current,  
           [0048]    generating a compensation signal which is approximated in its curve profile to the interference signal and has an amplitude and frequency which are identical to the interference signal but phase-shifted through 180°, wherein the compensation signal is subtracted from the differential signal and as a result largely cancels out the interference signal. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0049]    Exemplary embodiments according to the invention are explained in more detail below with reference to a schematic drawing in which:  
         [0050]    [0050]FIG. 1 shows a basic circuit of a linear lambda probe,  
         [0051]    [0051]FIG. 2 shows a known circuit arrangement for operating a linear lambda probe,  
         [0052]    [0052]FIG. 3 shows temperature dependence and pressure dependence of the transmission ratio of a linear lambda probe,  
         [0053]    [0053]FIG. 4 shows temperature dependence and equivalent circuit diagram of the probe impedance of a linear lambda probe,  
         [0054]    [0054]FIG. 5 shows a basic circuit arrangement for operating a linear lambda probe with a compensation circuit according to the invention,  
         [0055]    [0055]FIG. 6 shows a typical curve profile of the interference signal,  
         [0056]    [0056]FIG. 7 shows the alternating influence of the interference signal and of the pumping current,  
         [0057]    [0057]FIG. 8 shows a circuit diagram of a first exemplary embodiment of a compensation circuit,  
         [0058]    [0058]FIG. 9 shows an interference signal, compensation signal and residual signal of the circuit according to FIG. 8,  
         [0059]    [0059]FIG. 10 shows the integrator from the compensation circuit according to FIG. 8,  
         [0060]    [0060]FIG. 11 shows the output signal of the integrator with the signal Phi1 switched off and the actuation signal constant at 1.75 V,  
         [0061]    [0061]FIG. 12 shows the output signal of the integrator with the signal Phi1 switched on and the actuation signal constant at 1.75 V,  
         [0062]    [0062]FIG. 13 shows the output signal of the integrator with the signal Phi1 switched on and the actuation signal 1.5 kHz,  
         [0063]    [0063]FIG. 14 shows the output signal of the integrator with the signal Phi1 switched on and the actuation signal 6 kHz,  
         [0064]    [0064]FIG. 15 shows the output signal of the integrator with the signal Phi1 switched on and the actuation signal 3 kHz-90° phase-shifted,  
         [0065]    [0065]FIG. 16 shows the output signal of the integrator with the signal Phi1 switched on and the actuation signal 3 kHz without a phase shift,  
         [0066]    [0066]FIG. 17 shows a two-stage compensation circuit,  
         [0067]    [0067]FIG. 18 shows the signal profiles for the two-stage compensation, and  
         [0068]    [0068]FIG. 19 shows a compensation circuit which is integrated into the controller. 
     
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0069]    The invention is based on the idea that the amplitude of the interference signal (FIG. 6) is not known but the basic curve profile (square wave with time constant), the frequency and the phase angle are, as the generating signal originates of course from the local oscillator OSZ, V 12 , V 15 . If a further square wave (compensation signal) with the same amplitude and frequency with a 180° shifted phase is subtracted from this interference signal, as illustrated in FIG. 9, the two signals largely cancel one another out (filtering in the time domain).  
         [0070]    The magnitude of the residual signal is determined by the following factors:  
         [0071]    the phase difference between the interference signal and compensation signal (if both signals are generated locally, the phase difference may be negligibly small),  
         [0072]    amplitude difference between the two signals (this is minimized with the circuit described further below),  
         [0073]    differences in the signal profile (if the interference signal and compensation signal have different curve shapes, for example square wave with time constant and pure square wave, the time constant is retained in the residual signal but can be further reduced by incrementally approximating the curve profiles.  
         [0074]    The interference signal and the compensation signal (FIG. 9 above and center) have amplitudes of approximately 200 mV (ss), the residual signal (FIG. 9 below) only has an amplitude of 30 mV (ss) after slight low-pass filtering (τ=20 μs). The interference signal has therefore attenuated by approximately 16 dB.  
         [0075]    [0075]FIG. 8 shows a circuit KS according to the invention for compensating interference signals in the control loop of a linear lambda probe. In order to represent the behavior of the compensation circuit correctly, it is considered in conjunction with a probe model SM, the controller known from FIG. 5 and the 3 kHz oscillator V 12  and described in more detail with respect to its function.  
         [0076]    The probe model SM generates here a signal which comes as close as possible to the real interference signal as illustrated in FIG. 6. For this purpose, the signal source V 12  firstly generates a square wave signal Phi1 with a frequency of 3 kHz and 0 V/5 V levels. This signal is then attenuated to, for example, 200 mV (ss) with 100 mV offset by means of a voltage divider R 1 /Ri. A further signal source Vs additionally generates an offset of, for example, 2.15 V. In this way, a square wave signal of 200 mV (ss) with an offset of 2.25 V, which corresponds to the conditions with a real lambda probe, is obtained at its output.  
         [0077]    The square wave signal then passes to a filter network R 14   a,  R 14   b,  R 14   c,  C 14  and C 5  which gives the square wave signal the desired curve form (square wave with time constant).  
         [0078]    An amplifier AMP 5   b  forms, together with resistors R 14   a,  R 14   b  and R 304 , an inverter so that the 3 kHz signal ΔVs which is then filtered is produced with an amplitude of 200 mV (ss) at its output. Its noninverting input is at the center voltage Vm (2.25 V).  
         [0079]    The compensation circuit KS itself is composed of two amplifiers AMP 4   a  and AMP 5   a , four CMOS switches S 3   a ,S 3   b , S 3   c , S 3   d , three resistors R 22 , R 32 , R 302 , and of two capacitors C 50  and C 51 .  
         [0080]    The inverting input of AMP 4   a , which acts as a buffer, is connected to its output so that it forms an amplifier with a gain factor of 1. (By inserting two resistors (not illustrated) between the inverting input and output or inverting input and Vm it is also possible to set the circuit to higher gain factors.  
         [0081]    The noninverting input of AMP 4   a  is connected via a resistor R 302  to the output of AMP 5   b  and via a resistor R 22  to the output of the amplifier AMP 5   a . The noninverting input of AMP 5   a  is at Vm. The inverting input is connected via a series circuit composed of C 51  and R 2  to the output of AMP 4   a  as well as to the inputs of the switches S 3   b  and S 3   d . The output of AMP 5   a  is additionally connected to the inputs of the switches S 3   a  and S 3   c.    
         [0082]    The capacitor C 50  is connected between the outputs of the switches S 3   a -S 3   b  and the outputs of the switches S 3   c -S 3   d . The control inputs of S 3   a  and S 3   d  are connected to the signal source V 12 , and the control inputs of S 3   b  and S 3   d  are connected to Phi1 via one inverter (74HC04) each.  
         [0083]    The controller R is connected downstream of this compensation circuit KS. The embodiment shown represents an integral controller which filters relatively high frequency components more strongly. It is composed of an amplifier AMP 4   b , resistors R 41 , R 42  and a capacitor C 12  . The noninverting input of AMP 4   b  is connected to the center voltage Vm. The inverting input is connected via R 41  to the output of AMP 4   a , and via a parallel circuit composed of R 42  and C 12  to the output of AMP 4   a . R 42  is an equivalent resistor for the process of simulation, and it represents the finite amplification of AMP 4   b . It is not present in the real operation.  
         [0084]    The core of the compensation circuit KS is the integrator which is composed of AMP 5   a , C 50  and R 32 . The switches S 3   a , S 3   b , S 3   c , S 3   d  form, together with the inverters 74HC04, an alternating switch. The latter periodically reverses the polarity of the capacitor C 50  in synchronism with the oscillator signal Phi1 (3 kHz) between the inverting input of AMP 5   a  and its output so that a voltage which is integrated at the capacitor C 50  appears at the output of the integrator AMP 5   a  as a square wave signal.  
         [0085]    Switching over using the oscillator signal Phi1 also ensures that the integrator AMP 5   a  integrates only signal components of ΔVs which are phase-synchronous with respect to it. All other signal components are averaged out.  
         [0086]    FIGS.  10  to  16  and the description are used to illustrate the mode of operation of the integrator Amp 5   a.    
         [0087]    The integrator illustrated once more in FIG. 10 is viewed without the signal feedback through the resistor R 22 . R 302  and Amp 4   a  are also omitted so that the actuation takes place directly at the capacitor C 51 , which is additionally bridged (dashed line) in order to be able to bring about the behavior in the case of DC voltage actuation which is shown in FIG. 11.  
         [0088]    When the oscillator signal (Phi1-0V) is switched off, top of FIG. 11, the actuation signal Vin is a DC voltage with 1.75 V, in the center of FIG. 11. As the reference point of the integrator is at Vm=2.25 V (voltage at the noninverting input of Amp 5   a ), the actuation voltage and its output voltage must also be referred to Vm.  
         [0089]    The value of the actuation voltage referred to Vm is therefore ΔVin=1.75 V-Vref=−0.5 V. It is integrated with the time constant τ=R 32 *C 50 . After the time T, the output voltage of the integrator reaches the value: ΔVout={−ΔVin*T/τ}. Using the values selected in the example of ΔVin =−0.5 V, T=10 ms, R 32 =10 kOhm and C 51 =0.1 μF the following is obtained: ΔVout=5 V.  
         [0090]    If the oscillator signal Phi1 is then connected into the circuit (top of FIG. 12), the integrating capacitor C 50  is periodically switched over between the input and output of the integrator using the oscillator signal Phi1: each period of the signal ΔVs being integrated at the integrator Amp 5   a  takes place in two phases.  
         [0091]    During the first phase (Phi1=0 V), ΔVin is integrated to approximately 2.34 V (Vm+0.09 V) at the output of Amp 5   a  (bottom of FIG. 12).  
         [0092]    At the start of the second phase (Phi1=5 V), the capacitor C 50  is switched over and Vout jumps to approximately 2.17 V (Vm−0.08 V). Then (in the following signal period), integration is performed again in the following first phase up to 2.34 V etc.  
         [0093]    On average, the output voltage therefore remains stable at 2.25 V despite the DC input voltage. ΔVout=(2.34 V−2.17 V)=0.17 V.  
         [0094]    If the frequency of the actuation signal is changed, for example, to 1.5 kHz (center of FIG. 13) or 6 kHz (center of FIG. 14) in comparison with the oscillator signal Phi1 (top of FIG. 13, top of FIG. 14), only a relatively small, almost constant signal is also produced at the output of Amp 5   a . The cause of this is that the integrator averages over less (1.5 kHz) or more (6 kHz) than half a period of the actuation signal before it switches over. Correspondingly, the averaged residual amplitudes ΔVout (1.5 kHz) are at approximately 184 mV (bottom of FIG. 13) and ΔVout (6 kHz) at approximately 60 mV (bottom of FIG. 14). For AC voltage signals, the gain drops monotonously with 20 dB per decade of frequency increase.  
         [0095]    If the phase angle between Phi1 (top of FIG. 15) and the actuation signal is changed via, for example, 90° (center of FIG. 15) while the frequency is the same, a residual amplitude ΔVout of 102 mV (bottom of FIG. 15) is obtained. The actuation signal is also averaged out here.  
         [0096]    Only if Phi1 and the actuation signal have the same frequency and phase angle, as illustrated at the top and in the center of FIG. 16, is a continuously increasing signal obtained at the integrator output (bottom of FIG. 16). The integration phase and signal phase have identical profiles so that averaging is not carried out but instead integration, which is analogous to actuating with DC voltage.  
         [0097]    Since the actuation signal Vin and the integrator have the same reference potential (2.25 V), the connection can be made by means of the capacitor C 51 . In this way, the influence of any control error of the pumping current control (ΔVs) is reduced without disadvantages.  
         [0098]    The function of the integrator in the real compensation circuit (FIGS. 5 and 8) is explained from the behavior of the integrator as described in FIGS.  10  to  16 .  
         [0099]    As long as the actuation signal Vin does not have any signal components which are frequency-synchronous and phase-synchronous with the oscillator signal Phi1, the output of the integrator Amp 5   a  will only have a DC voltage (with Vm=2.25 V). The output impedance of Amp 5   a  is small here. The signal ΔVs passes via the resistor R 302  (FIG. 8) to the noninverting input of Amp 4   a , and it is attenuated by the voltage dividers R 302 , R 22 . If the resistors R 302  and R 22  have the same value, the attenuation is 50%.  
         [0100]    From the output of the amplifier Amp 4   a , said signal passes onto the input of the controller R. Overall, the compensation signal for ΔVs therefore only has the effect of a voltage divider. However, this is without significance as the amplitude loss can be compensated by a suitable configuration of the controller R.  
         [0101]    However, if the actuation signal Vin has signal components which are frequency-synchronous and phase-synchronous with the oscillator signal Phi1, for example, the 3 kHz signal, acting as interference signal, of the Ris measurement (FIG. 6), this component is integrated at the integrator Amp 5   a . A 3 kHz square wave signal with rising amplitude and phase-shifted with respect to the interference signal contained in the differential signal ΔVs by 180° appears at the output of Amp 5   a .  
         [0102]    As a consequence of this output signal, the 3 kHz amplitude at the noninverting input of Amp 4   a  is reduced. Correspondingly, the 3 kHz amplitude at the output of Amp 4   a  also drops so that the actuation signal of the integrator Amp 5   a  is also reduced.  
         [0103]    Ultimately, a state of equilibrium is established, the interference signal component in ΔVs and the compensation signal (3 kHz output signal of the integrator Amp 5   a ) being largely cancelled out so that only a residual signal remains at the integrator input and thus at the output of the compensation circuit. On the other hand, all the other frequency components of ΔVs are only attenuated by the 50% described above.  
         [0104]    The control loop which is produced in this way compensates the interference signal completely if the curve shape is a rectangle. However, as this is not the case in practice, the compensation can be improved by approximating it incrementally to the real curve profile.  
         [0105]    An expansion to multi-step characteristics (a plurality of integration phase sections per signal period) is carried out by adding at least one further capacitor and further changeover switches to the integrator and incrementally carrying out the integration of the 3 kHz signal using these capacitors. A corresponding circuit is illustrated in FIG. 17.  
         [0106]    A further capacitor C 52  is inserted, one terminal of which capacitor C 52  is connected to the outputs of the switches S 3   a , S 3   b , in parallel with the capacitor C 50 . The other terminal of C 50  is then no longer connected to the outputs of the switches S 3   c , S 3   d , but rather to the input of a switch S 3   f . The other terminal of C 52  is likewise connected to the input of a switch S 3   e . The outputs of S33 and S 3   f  are connected to the outputs of S 3   c , S 3   d . The control input of S 3   e  is connected to a further signal source V 15  which generates an oscillator signal Phi2 with a frequency of 6 kHz. The control input of S 3   f  is also connected to the signal source V 15  via a further inverter (74HC04). The oscillator signal (Phi1=3 kHz) can be generated, for example, by halving the 6 kHz oscillator signal (Phi2) by means of a frequency divider (FIG. 17).  
         [0107]    By means of this expansion, the integration of each period of the signal ΔVs at the integrator Amp 5   a  is then decomposed into 4 phase sections:  
         [0108]    In the phase section 1 (0%-25% of the period length of the oscillator Phi1) it will be assumed that capacitor C 50  is connected to the other circuit via switch S 3   f . Said capacitor C 50  is therefore active as an integrating capacitor. Due to the control effect of the compensation circuit KS, a phase-synchronous amplitude value which corresponds to the value (low at this time owing to the time constant) of the interference signal is produced at the output of AMP 5   a.    
         [0109]    In phase section 2 (25%-50%), switch S 3   f  is opened and switch S 3   e  is closed as a result of the level change of the signal source V 15 . Now, capacitor C 52  is active as the integrating capacitor. It then integrates the (risen) value of the interference signal. Correspondingly, the output voltage of the integrator is somewhat higher in this phase.  
         [0110]    Phase section 3 (50%-75%) corresponds to phase section 1, but now, due to the position of the alternating switches, the capacitor C 52  is active as an integrating capacitor and the amplitude of the integrator output jumps from positive to negative.  
         [0111]    Phase section 4 (75%-100%) corresponds in turn to phase section 2 (C 52  active), the amplitude being also negative here.  
         [0112]    The switch positions in the individual phases can be found in the following table.  
                                                                                                 S3a   S3b   S3c   S3d   S3e   S3f                                    Phase section 1   On   Off   Off   On   Off   On       Phase section 2   On   Off   Off   On   On   Off       Phase section 3   Off   On   On   Off   Off   On       Phase section 4   Off   On   On   Off   On   Off                  
 
         [0113]    Thus, the curve profile of the compensation signal illustrated in FIG. 18, track 4 is obtained. As a result of this, the (filtered) residual signal is reduced from approximately 30 mV (ss) for single-switch compensation to approximately 13 mV (ss) for dual-switch compensation, that is to say via further 7 dB. Overall, the interference signal is then attenuated by 16dB+7dB=23 dB.  
         [0114]    [0114]FIG. 18 shows the following signals:  
                                       Track 1:   oscillator signal Phi1 = 3 kHz,       Track 2:   oscillator signal Phi2 = 6 kHz,       Track 3:   interference signal contained in the differential signal ΔVs,       Track 4:   compensation signal of two-stage composition           (at the output of the integrator Amp5a),       Track 5:   residual signal at the output of the compensation circuit,       and       Track 6:   residual signal at the output of the controller after further           filtering.                  
 
         [0115]    A further subdivision of the integration intervals leads to an increase in the improvement in this attenuation, but also to greater expenditure on hardware and software.  
         [0116]    In addition to the implementational example presented, alternative embodiments of the invention are also conceivable.  
         [0117]    For example, the compensation circuit can be combined with the controller R. In FIG. 19, the integration of the compensation circuit according to FIG. 8 is combined with the controller R, as a result of which the amplitude loss, due to the voltage dividers R 302 , R 22  from FIG. 8, is avoided.  
         [0118]    In contrast with the circuit illustrated in FIGS. 8 and 17, the amplifier Amp 4   a  is not operated here as a buffer but rather as an inverter by means of the resistors R 22  and R 302 . Its noninverting input is at the center voltage Vm (2.25 V). The output of the inverter Amp 4   a  constitutes the output of the compensation circuit and is connected via a resistor R 43  to the inverting input of the controller Amp 4   b , in the same way as R 41  has already been connected. As a result, the controller is expanded to form a summing element. The output of the controller is connected to the capacitor C 51 , the input of the compensation circuit. The compensation effect then takes place as a result of the summing property of the controller R.  
         [0119]    The invention as described can be used not only in a circuit arrangement for compensating interference signals in the control loop of a linear lambda probe but also quite generally in control circuits for compensating interference variables.