Abstract:
A circuit coupled to receive a sequence of signals is designed with a multiplication circuit ( 416, 420 ) coupled to receive a first signal, a second signal and a complex conjugate of the first signal. The second signal follows the first signal in time. The multiplication circuit produces a first product sequence of the first signal and the complex conjugate and a second product sequence of the second signal and the complex conjugate. A summation circuit ( 424, 426 ) is coupled to receive the first product sequence and the second product sequence. The summation circuit produces a first sum of the first product sequence and a second sum of the second product sequence.

Description:
FIELD OF THE INVENTION 
     This invention relates to wideband code division multiple access (WCDMA) for a communication system and more particularly to Doppler frequency estimation of WCDMA signals with space-time transmit diversity. 
     BACKGROUND OF THE INVENTION 
     Present code division multiple access (CDMA) systems are characterized by simultaneous transmission of different data signals over a common channel by assigning each signal a unique code. This unique code is matched with a code of a selected receiver to determine the proper recipient of a data signal. These different data signals arrive at the receiver via multiple paths due to ground clutter and unpredictable signal reflection. Additive effects of these multiple data signals at the receiver may result in significant fading or variation in received signal strength. In general, this fading due to multiple data paths may be diminished by spreading the transmitted energy over a wide bandwidth. This wide bandwidth results in greatly reduced fading compared to narrow band transmission modes such as frequency division multiple access (FDMA) or time division multiple access (TDMA). 
     New standards are continually emerging for next generation wideband code division multiple access (WCDMA) communication systems as described in Provisional U.S. Patent Application No. 60/082,671, filed Apr. 22, 1998, and incorporated herein by reference. These WCDMA systems are coherent communications systems with pilot symbol assisted channel estimation schemes. These pilot symbols are transmitted as quadrature phase shift keyed (QPSK) known data in predetermined time frames to any receivers within range. The frames may propagate in a discontinuous transmission (DTX) mode. For voice traffic, transmission of user data occurs when the user speaks, but no data symbol transmission occurs when the user is silent. Similarly for packet data, the user data may be transmitted only when packets are ready to be sent. The frames are subdivided into sixteen equal time slots of 0.625 milliseconds each. Each time slot is further subdivided into equal symbol times. At a data rate of 32 KSPS, for example, each time slot includes twenty symbol times. Each frame includes pilot symbols as well as other control symbols such as transmit power control (TPC) symbols and rate information (RI) symbols. These control symbols include multiple bits otherwise known as chips to distinguish them from data bits. The chip transmission time (T c ), therefore, is equal to the symbol time rate (T) divided by the number of chips in the symbol (N). 
     Previous studies have shown that multiple transmit antennas may improve reception by increasing transmit diversity for narrow band communication systems. In their paper  New Detection Schemes for Transmit Diversity with no Channel Estimation,  Tarokh et al. describe such a transmit diversity scheme for a TDMA system. The same concept is described in  A Simple Transmitter Diversity Technique for Wireless Communications  by Alamouti. Tarokh et al. and Alamouti, however, fail to teach such a transmit diversity scheme for a WCDMA communication system. 
     Other studies have investigated open loop transmit diversity schemes such as orthogonal transmit diversity (OTD) and time switched time diversity (TSTD) for WCDMA systems. Both OTD and TSTD systems have similar performance. Both use multiple transmit antennas to provide some diversity against fading, particularly at low Doppler rates and when there are insufficient paths for the rake receiver. Both OTD and TSTD systems, however, fail to exploit the extra path diversity that is possible for open loop systems. For example, the OTD encoder circuit of FIG. 5 receives symbols S 1 , and S 2  on lead  500  and produces output signals on leads  504  and  506  for transmission by first and second antennas, respectively. These transmitted signals are received by a despreader input circuit (not shown). The despreader circuit sums received chip signals over a respective symbol time to produce first and second output signals R j   1  and R j   2  on leads  620  and  622  as in equations [1-2], respectively.                R   j   1     =         ∑     i   =   0       N   -   1                         r   j          (     i   +     τ   j       )         =         α   j   1          S   1       +       α   j   2          S   2                   [   1   ]                 R   j   2     =         ∑     i   =   N         2      N     -   1                         r   j          (     i   +     τ   j       )         =         α   j   1          S   1       -       α   j   2          S   2                   [   2   ]                                
     The OTD phase correction circuit of FIG. 6 receives the output signals R j   1  and R j   2  corresponding to the j th  of L multiple signal paths. The phase correction circuit produces soft outputs or signal estimates {tilde over (S)} 1  and {tilde over (S)} 2  for symbols S 1  and S 2  at leads  616  and  618  as shown in equations [3-4], respectively.                  S   ~     1     =         ∑     j   =   1     L                       (       R   j   1     +     R   j   2       )          α   j     1   *           =       ∑     j   =   1     L                     2               α   j   1          2          S   1                   [   3   ]                   S   ~     2     =         ∑     j   =   1     L                       (       R   j   1     -     R   j   2       )          α   j     2   *           =       ∑     j   =   1     L                     2               α   j   2          2          S   2                   [   4   ]                                
     Equations [3-4] show that the OTD method provides a single channel estimate α for each path j. A similar analysis for the TSTD system yields the same result. The OTD and TSTD methods, therefore, are limited to a path diversity of L. This path diversity limitation fails to exploit the extra path diversity that is possible for open loop systems as will be explained in detail. 
     Hosur et al. previously taught a new method for frame synchronization with space time transmit diversity (STTD) having a path diversity of 2L in U.S. Pat. application Ser. No. 09/195,942, filed Nov. 19, 1998, and incorporated herein by reference. Therein, Hosur et al. taught advantages of this increased diversity for WCDMA systems. Hosur et al. did not teach or suggest how this improved diversity might relate to Doppler frequency estimation. 
     Doppler frequency estimation is particularly critical in WCDMA systems where a mobile receiver may move with respect to a base station by car or high-speed train. Such motion may produce an apparent change of frequency or Doppler frequency shift of over 500 Hz. A reliable Doppler frequency estimate is important for Rayleigh fading parameter channel estimates, transmit power control (TPC) estimates and efficient use of downlink transmit antenna diversity such as STTD. Knowledge of the Doppler frequency is equally important for other channel estimate schemes such as iterative channel estimation (ICE) or weighted multi-slot averaging (WMSA). For example, knowledge of a Doppler frequency shift permits use of an optimal Weiner filter for channel estimates. If the Doppler frequency shift is unknown, the same filter must be used for a wide range of Doppler frequencies resulting in a degraded link margin. Use of an optimal filter is also highly advantageous in TPC estimation. Furthermore, downlink transmit antenna diversity operates differently with varying Doppler frequencies. At low Doppler frequencies, for example, a mobile system may send information to the base station indicating which antenna signal is stronger. At high Doppler frequencies, space-time code across transmit antennas may be used to achieve transmit diversity. 
     SUMMARY OF THE INVENTION 
     The foregoing problems are resolved by a circuit coupled to receive a sequence of signals comprising a multiplication circuit coupled to receive a first signal, a second signal and a complex conjugate of the first signal. The second signal follows the first signal in time. The multiplication circuit produces a first product sequence of the first signal and the complex conjugate and a second product sequence of the second signal and the complex conjugate. A summation circuit is coupled to receive the first product sequence and the second product sequence. The summation circuit produces a first sum of the first product sequence and a second sum of the second product sequence. 
     The present invention provides highly accurate estimates of Doppler frequency shift. The estimates are compatible with STTD and other diversity schemes. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     A more complete understanding of the invention may be gained by reading the subsequent detailed description with reference to the drawings wherein: 
     FIG. 1 is a simplified block diagram of a typical transmitter using Space Time Transit Diversity (STTD) of the present invention; 
     FIG. 2 is a block diagram showing signal flow in an STTD encoder of the present invention that may be used with the transmitter of FIG. 1; 
     FIG. 3 is a schematic diagram of a phase correction circuit of the present invention that may be used with a receiver; 
     FIG. 4 is a schematic diagram of a circuit for calculating autocorrelation values; 
     FIG. 5 is a block diagram showing signal flow in an OTD encoder of the prior art; 
     FIG. 6 is a schematic diagram of a phase correction circuit of the prior art; and 
     FIG. 7 is a plot of autocorrelation values for various Doppler frequencies over multiple time slots. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Referring to FIG. 1, there is a simplified block diagram of a typical transmitter using Space Time Transit Diversity (STTD) of the present invention. The transmitter circuit receives pilot symbols, TPC symbols, RI symbols and data symbols on leads  100 ,  102 ,  104  and  106 , respectively. Each of the symbols is encoded by a respective STTD encoder as will be explained in detail. Each STTD encoder produces two output signals that are applied to multiplex circuit  120 . The multiplex circuit  120  produces each encoded symbol in a respective symbol time of a frame. Thus, a serial sequence of symbols in each frame is simultaneously applied to each respective multiplier circuit  124  and  126 . A channel orthogonal code C m  is multiplied by each symbol to provide a unique signal for a designated receiver. The STTD encoded frames are then applied to antennas  128  and  130  for transmission. 
     Turning now to FIG. 2, there is a block diagram showing signal flow in an STTD encoder of the present invention that may be used with the transmitter of FIG. 1 for pilot symbol encoding. The pilot symbols are predetermined control signals that may be used for channel estimation and other functions as will be described in detail. Operation of the STTD encoder  112  will be explained with reference to TABLE 1. The STTD encoder receives pilot symbol  11  at symbol time T, pilot symbol S 1  at symbol time  2 T, pilot symbol  11  at symbol time  3 T and pilot symbol S 2  at symbol time  4 T on lead  100  for each of sixteen time slots of a frame. For a first embodiment of the present invention having a data rate of preferably 32 KSPS, the STTD encoder produces a sequence of four pilot symbols for each of two antennas corresponding to leads  204  and  206 , respectively, for each of the sixteen time slots of TABLE 1. The STTD encoder produces pilot symbols B 1 , S 1 , B 2  and S 2  at symbol times T- 4 T, respectively, for a first antenna at lead  204 . The STTD encoder simultaneously produces pilot symbols B 1 , −S* 2 , −B 2  and S* 1  at symbol times T- 4 T, respectively, at lead  206  for a second antenna. Each symbol includes two bits representing a real and imaginary component. An asterisk indicates a complex conjugate operation or sign change of the imaginary part of the symbol. Pilot symbol values for the first time slot for the first antenna at lead  204 , therefore, are  11 ,  11 ,  11  and  11 . Corresponding pilot symbols for the second antenna at lead  206  are  11 ,  01 ,  00 , and  10 . 
     The bit signals r j (i+τ j ) of these symbols are transmitted serially along respective paths  208  and  210 . Each bit signal of a respective symbol is subsequently received at a remote mobile antenna  212  after a transmit time τ corresponding to the j th  path. The signals propagate to a despreader input circuit (not shown) where they are summed over each respective symbol time to produce input signals R j   1 , R j   2 , R j   3  and R j   4  corresponding to the four pilot symbol time slots and the j th  of L multiple signal paths as previously described. 
     
       
         
               
               
               
             
               
               
               
               
               
               
               
               
               
             
               
               
               
               
               
               
               
               
               
             
           
               
                   
                 TABLE 1 
               
             
             
               
                   
                   
               
               
                   
                 ANTENNA 1 
                 ANTENNA 2 
               
             
          
           
               
                 SLOT 
                 B 1   
                 S 1   
                 B 2   
                 S 2   
                 B 1   
                 −S* 2   
                 −B 2   
                 S* 1   
               
               
                   
               
             
          
           
               
                 1 
                 11 
                 11 
                 11 
                 11 
                 11 
                 01 
                 00 
                 10 
               
               
                 2 
                 11 
                 11 
                 11 
                 01 
                 11 
                 11 
                 00 
                 10 
               
               
                 3 
                 11 
                 01 
                 11 
                 01 
                 11 
                 11 
                 00 
                 00 
               
               
                 4 
                 11 
                 10 
                 11 
                 01 
                 11 
                 11 
                 00 
                 11 
               
               
                 5 
                 11 
                 10 
                 11 
                 11 
                 11 
                 01 
                 00 
                 11 
               
               
                 6 
                 11 
                 10 
                 11 
                 11 
                 11 
                 01 
                 00 
                 11 
               
               
                 7 
                 11 
                 01 
                 11 
                 00 
                 11 
                 10 
                 00 
                 00 
               
               
                 8 
                 11 
                 10 
                 11 
                 01 
                 11 
                 11 
                 00 
                 11 
               
               
                 9 
                 11 
                 11 
                 11 
                 00 
                 11 
                 10 
                 00 
                 10 
               
               
                 10 
                 11 
                 01 
                 11 
                 01 
                 11 
                 11 
                 00 
                 00 
               
               
                 11 
                 11 
                 11 
                 11 
                 10 
                 11 
                 00 
                 00 
                 10 
               
               
                 12 
                 11 
                 01 
                 11 
                 01 
                 11 
                 11 
                 00 
                 00 
               
               
                 13 
                 11 
                 00 
                 11 
                 01 
                 11 
                 11 
                 00 
                 01 
               
               
                 14 
                 11 
                 10 
                 11 
                 00 
                 11 
                 10 
                 00 
                 11 
               
               
                 15 
                 11 
                 01 
                 11 
                 00 
                 11 
                 10 
                 00 
                 00 
               
               
                 16 
                 11 
                 00 
                 11 
                 00 
                 11 
                 10 
                 00 
                 01 
               
               
                   
               
             
          
         
       
     
     The input signals corresponding to the pilot symbols for each time slot are given in equations [5-8]. Noise terms are omitted for simplicity. Received signal R j   1  is produced by pilot symbols (B 1 ,B 1 ) constant value (11,11) at symbol time T for all time slots. Thus, the the sum of respective Rayleigh fading parameters corresponding to the first and second antennas. Likewise, received signal R j   3  is produced by pilot symbols (B 2 ,−B 2 ) having a constant value (11,00) at symbol time  3 T for all time slots. Channel estimates for the Rayleigh fading parameters corresponding to the first and second antennas, therefore, are readily obtained from input signals R j   1  and R j   3  as in equations [9] and [10]. 
     
       
           R   j   1 =α j   1 +α j   2   [5] 
       
     
     
       
           R   j   2 =α j   1   S   1 −α j   2   S*   2   [6] 
       
     
     
       
           R   j   3 =α j   1 −α j   2   [7] 
       
     
     
       
           R   j   4 =α j   1   S   2 +α j   2   S*   1   [8] 
       
     
     
       
         α j   1 =( R   j   1   +R   j   3 )/2  [9] 
       
     
     
       
         α j   2 =( R   j   1   −R   j   3 )/2  [10] 
       
     
     Referring now to FIG. 3, there is a schematic diagram of a phase correction circuit of the present invention that may be used with a remote mobile receiver. This phase correction circuit receives input signals R j   2  and R j   4  on leads  324  and  326  at symbol times  2 T and  4 T, respectively. Each input signal has a value determined by the transmitted pilot symbols as shown in equations [6] and [8], respectively. The phase correction circuit receives a complex conjugate of a channel estimate of a Rayleigh fading parameter α j   1 * corresponding to the first antenna on lead  302  and a channel estimate of another Rayleigh fading parameter α j   2  corresponding to the second antenna on lead  306 . Complex conjugates of the input signals are produced by circuits  308  and  330  at leads  310  and  322 , respectively. These input signals and their complex conjugates are multiplied by Rayleigh fading parameter estimate signals and summed as indicated to produce path-specific first and second symbol estimates at respective output leads  318  and  322  as in equations [11] and [12]. 
     
       
           R   j   2 α j   1   *+R   j   4 *α j   2 =(|α j   1 | 2 +|α j   2 | 2 ) S   1   [11] 
       
     
     
       
         − R   j   2 *α j   2   +R   j   4 α j   1 *=(|α j   1 | 2 +|α j   2 | 2 ) S   2   [12] 
       
     
     These path-specific symbol estimates are then applied to a rake combiner circuit (not shown) to sum individual path-specific symbol estimates, thereby providing net soft symbols or pilot symbol signals as in equations [13] and [14].                  S   ~     1     =         ∑     j   =   1     L                       R   j   2          α   j     1   *           +       R   j     4   *            α   j   2                 [   13   ]                   S   ~     2     =         ∑     j   =   1     L                       -     R   j     2   *              α   j   2         +       R   j   4          α   j     1   *                   [   14   ]                                
     These soft symbols or estimates provide a path diversity L and a transmit diversity 2. Thus, the total diversity of the STTD system is 2L. This increased diversity is highly advantageous in providing a reduced bit error rate. 
     Referring now to FIG. 4, there is a schematic diagram of a circuit for calculating autocorrelation values with STTD. The soft symbols or pilot symbol signals from the rake combiner as in equations [13-14] are summed for each respective time slot. A sequence of these summed pilot symbol signals corresponding to each respective time slot is applied to lead  400 . An exemplary summed pilot symbol signal, for example p k+2  corresponding to a sum of pilot symbol signals of time slot k+2, is applied to conjugate circuit  402 . Conjugate circuit  402  inverts the imaginary component of the summed pilot symbol signal to produce conjugate signal p* k+2  on lead  404 . Delay circuit  406  produces delayed pilot symbol signal p* k  from the k th  time slot on lead  408 . Multiplier circuit  410  applies a product p* k p k+   2  on lead  440  to summation circuit  422 . Summation circuit  422  accumulates each product of the sequence for each time slot for preferably one second and produces autocorrelation value C 2  as in equation [17]. Delay circuit  412  produces delayed summed pilot symbol signal p k+1  from time slot k+1 on lead  414 . Multiplier circuit  416  produces a product p* k p k+1  on lead  442 . Summation circuit  424  accumulates each product on lead  442  for preferably one second and produces autocorrelation value C 1  as in equation [16]. Delay circuit  418  produces delayed summed pilot symbol signal p k  on lead  434 . Multiplier circuit  420  produces a product p* k p k  on lead  444 , and summation circuit  426  accumulates each product for preferably one second and produces autocorrelation value C 0  as in equation [15].                C   0     =       ∑     k   =   1     1600                       p   k   *          p   k                 [   15   ]                 C   1     =       ∑     k   =   1     1600                       p   k   *          p     k   +   1                   [   16   ]                 C   2     =       ∑     k   =   1     1600                       p   k   *          p     k   +   2                   [   17   ]                                
     Autocorrelation values C 1  and C 2  are divided by C 0  for normalization. Noise and interference create an impulse for zero delay of autocorrelation value C 0 . This impulse effectively multiplies C 0  by            S   +   I     S     ,                          
     where S is signal power and I is noise plus interference power. Thus, normalized autocorrelation values have a form of equations [18-19].                  C   ^     1     =         (     S   +   I     )     ·     C   1         S   ·     C   0                 [   18   ]                   C   ^     2     =         (     S   +   I     )     ·     C   2         S   ·     C   0                 [   19   ]                                
     Application of these normalized autocorrelation values to estimate Doppler frequencies will now be explained in detail with reference to FIG.  7 . The family of normalized autocorrelation curves of FIG. 7 are discrete points of Bessel functions corresponding to respective time slots. The curves correspond to respective Doppler frequencies of 100 Hz through 700 Hz in 100 Hz increments. Normalized autocorrelation values Ĉ 1  and Ĉ 2  lie along the vertical axes corresponding to time slots one and two. Known values of Bessel functions are used to estimate Doppler frequencies from the normalized autocorrelation values of equations. For example, if normalized autocorrelation value Ĉ 1  is greater than or equal to 0.5, then the Doppler frequency is close to the 100 Hz to 400 Hz curves. Alternatively, if normalized autocorrelation value Ĉ 1  is less than 0.5, then the Doppler frequency is close to the 400 Hz to 700 Hz curves. An optimal estimate of the Doppler frequency, therefore, is one that minimizes the mean squared error between the known Bessel function values and the normalized autocorrelation values as in equation [20]. These Bessel function values correspond to frequencies f k1  and f k2  for time slots  1  and  2 , respectively. 
     
       
         Doppler frequency index=arg min(| f   k1   −Ĉ   1 | 2   +|f   k2   −Ĉ   2 | 2 )  [20] 
       
     
     Doppler frequency estimation simulations were conducted to determine the accuracy of estimates for Doppler frequencies of 200 Hz, 400 Hz and 600 Hz. All estimates were correct for ten thousand trials at each frequency with a received bit energy-to-noise ratio (E 0 /N 0 ) of −3 dB. At a received bit energy to noise ratio (E 0 /N 0 ) of −10 dB, there were no errors at 200 Hz, one error at 400 Hz and one hundred twenty-six errors at 600 Hz. The one error at 400 Hz was a 300 Hz estimate. The 600 Hz errors included one hundred twenty-four 500 Hz estimates and two 700 Hz estimates. Thus, Doppler frequency estimates of the present invention are very accurate. Moreover, calculation of these estimates is relatively straightforward and typically requires about 0.25 MIPS. 
     Although the invention has been described in detail with reference to its preferred embodiment, it is to be understood that this description is by way of example only and is not to be construed in a limiting sense. For example, Doppler frequency estimation of the present invention may be extended to include additional autocorrelation values corresponding to other time slots. Additionally, autocorrelation values need not be normalized as in equations [18-19] to realize advantages of the present invention. Bessel function values corresponding to frequencies f k1  and f k2  may be multiplied by denominator S·C 0  with the same result in equation [19]. Moreover, Doppler frequency estimation need not be restricted to pilot symbol signals. Data symbols, TPC symbols and RI symbols in each respective time slot may be corrected by a phase correction circuit and used as virtual pilot symbols to enhance Doppler frequency estimation. Furthermore, advantages of the present invention are also achieved with other transmit diversity schemes such as time domain transmit diversity (FDTD) as disclosed in volume 3 of the  Association of Radio Industries and Businesses  (ARIB) specification (1998). Advantages of the present invention may also be achieved by a digital signal processing circuit as will be appreciated by those of ordinary skill in the art having access to the instant specification. 
     It is understood that the inventive concept of the present invention may be embodied in a mobile communication system as well as circuits within the mobile communication system. It is to be further understood that numerous changes in the details of the embodiments of the invention will be apparent to persons of ordinary skill in the art having reference to this description. It is contemplated that such changes and additional embodiments are within the spirit and true scope of the invention as claimed below.