Abstract:
A voltage regulator includes a voltage source for providing an input voltage and circuitry for regulating the input voltage to provide an output voltage. The circuitry for regulating the input voltage includes at least a high side switch and a low side switch. A skip mode controller controls the high side switch and the low side switch in order to minimize conduction losses and switching losses within the voltage regulator.

Description:
TECHNICAL FIELD OF THE INVENTION 
     The present invention relates to voltage regulators, and more particularly, to system and method for improving switching losses within a voltage regulator. 
     BACKGROUND OF THE INVENTION 
     A buck regulator is a switching power supply including at least one series switching transistor that chops the input voltage and applies the pulses to an average inductive capacitive filter. The output voltage of a buck regulator is lower than the input voltage. Buck regulators are one type of pulse width modulated (PWM) converters which are switching power supplies using power semiconductor switches in the on and off switching states to provide a device with high efficiency, small size and light weight. Pulse width modulated converters employ square wave pulse width modulation to achieve voltage regulation. The output voltage of the PWM converter is varied by varying the duty cycle of the power semiconductor switches within the circuit. The voltage waveform across the switches at the output is square wave in nature and generally results in higher switching losses when the switching frequency is increased. Traditional synchronous buck converters suffer from low light load efficiencies due to the high switching losses and high conduction losses created by the circuit. While circuitries have been developed for controlling the high conduction losses within traditional synchronous buck converters, there has been no design that provides improvement for both switching and conduction losses and provides for smooth transitions between the discontinuous current mode and continuous current mode of operation of the converter. Circuitry operating in this fashion would minimize power losses and thus improve the longevity of the power supply&#39;s operation. 
     SUMMARY OF THE INVENTION 
     The present invention overcomes the foregoing and other problems with a voltage regulator including the voltage source for providing an input voltage. The voltage regulator further includes associated circuitry for regulating the input voltage to provide an output voltage. The associated circuitry includes at least a high side switch and a low side switch. A skip mode controller controls the high side switch and low switch to minimize the conduction losses and the switching losses within the voltage regulator. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       For a more complete understanding of the present invention and the advantages thereof, reference is now made to the following description taken in conjunction with the accompanying Drawings in which: 
         FIG. 1  is a schematic diagram of a synchronous buck regulator having traditional constant frequency peak current mode control; 
         FIG. 2  is a timing diagram illustrating the operation of the circuit of  FIG. 1 ; 
         FIG. 3  is a schematic diagram of a synchronous buck regulator having zero crossing detection circuitry for limiting high conduction losses during light load operation; 
         FIG. 4  is a timing diagram illustrating the operation of the circuitry of  FIG. 3 ; 
         FIG. 5  is a schematic diagram of a synchronous buck regulator including a skip mode controller; 
         FIG. 6  is a schematic diagram of the skip mode controller; 
         FIG. 7  is a flow diagram illustrating the process for turning on the high side transistor of the synchronous buck regulator; 
         FIG. 8  is a flow diagram illustrating the process for turning off the high side transistor of the synchronous buck regulator; 
         FIG. 9  is a timing diagram illustrating the operation of the synchronous buck regulator of  FIG. 5 ; and 
         FIG. 10  is an illustration of the measured efficiencies of a regulator using the pulse skipping method of the present invention and regulator using a forced pulse width modulated approach. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Referring now to the drawings, and more particularly to  FIG. 1 , there is illustrated a synchronous buck generator  110  including traditional constant frequency peak current mode control. The high side switch  112  and low side switch  114  are controlled complementarily from the outputs of the RS flip flop  124 . A PWM comparator  116  compares the integrated voltage feedback signal VCOMP which is applied to the positive input of the PWM comparator  116  with the sum of the amplified current-sense signal from the current-sense amplifier  118  and a slope compensation ramp signal  120 . The output of the PWM comparator  116  is applied to RS flip-flop  124 , at each rising clock edge, the high side switch  112 , consisting of a MOSFET transistor, is turned on until the sum of the amplified current signal from the current signal amplifier  118  and the slope compensation signal  120  is greater than the integrated voltage feedback signal from the error amplifier  122 . When this signal condition is reached, the output of PWM comparator  116  resets the RS flip-flop  124  and the RS flip-flop  124  turns off the high side switch  112 . 
     Referring now also to  FIG. 2 , there is illustrated a timing diagram of the switching waveforms at light load in forced pulse width modulated (PWM) mode. While the high side transistor switch  112  is turned on, current I L  ramps up through the inductor  126  sourcing the current to the output (V OUT ) and storing energy within the inductor  126 . The current mode feedback system regulates the peak inductor current as a function of the output voltage error signal which is provided from the output of the error amplifier  122 . To preserve loop stability, a compensation ramp signal  120  is summed with the amplified current-sense signal from the current-sense amplifier  118 . When the high side switch  112  is turned off, the low side switch  114 , consisting of a MOSFET transistor, is turned on. The inductor  126  releases its stored energy as the current ramps down ( 204 ) in the off condition while still providing current to the output V OUT . The output capacitor  128  stores charge when the inductor  126  exceeds the load current. The output capacitor  128  releases this charge when the inductor current is lower and smooths the voltage across load  130 . 
     If the load current ( 206 ) is less than half of the peak inductor current ( 208 ), the inductor current I L  becomes negative in a certain amount of time interval ( 210 ) and circulates through the low side switch  114  resulting in high conduction losses. Switches  112  and  114  turn on and off complementarily with a fixed switching frequency responsive to the Q and  Q  outputs of the RS flip-flop  124 . Thus, when the valley inductor current value ( 212 ) is reached, the low side switch  114  will turn off and the high side switch  112  will turn on as illustrated at T 3 . Similarly, when the peak value ( 208 ) of the inductor current I L  is reached, the high side switch  112  turns off and the low side switch  114  turns on as illustrated at T 2 . The peak-to-peak inductor current (ΔI L ) may be determined by the equation; 
     
       
         
           
             
               Δ 
               ⁢ 
               
                   
               
               ⁢ 
               
                 I 
                 L 
               
             
             = 
             
               
                 
                   
                     V 
                     IN 
                   
                   - 
                   
                     V 
                     OUT 
                   
                 
                 L 
               
               · 
               
                 DT 
                 S 
               
             
           
         
       
     
     Referring now to  FIG. 3 , there is illustrated one prior art method for reducing the conduction losses within a synchronous buck regulator at light loads. The circuitry described in  FIG. 3  is the same as that described with respect to  FIG. 1  with the inclusion of the zero crossing detection circuitry  302 . The zero crossing detection circuitry  302  detects when the low side switch  114  current is below zero and turns off the low side switch  114  upon detection of this condition. 
     Referring now also to  FIG. 4 , there are illustrated the switching waveforms associated with the circuit of  FIG. 3 . The high side transistor  112  is turned on responsive to a clock pulse  402  at T 1 . The high side switch  112  remains on until the inductor current ramps up to its peak value ( 404 ). Upon reaching the peak inductor current value ( 404 ), the high side transistor  112  turns off and the low side transistor  114  turns on. The low side transistor  114  will remain on until point T 2  wherein the low side transistor  114  is turned off. The low side transistor  114  is turned off by the zero detection circuit  302  once the inductor current through the low side switch  114  drops below zero ( 406 ). Both the high side switch  112  and the low side switch  114  remain off between points T 2  and T 3  until receipt of the next clock pulse  406 . The process then repeats. There is no power loss while both switches  112  and  114  are turned off between T 2  and T 3 . This results in lower conduction losses when compared with the regulator circuit described with respect to  FIG. 1 . The output voltage V 0  is regulated by the duty cycle while the converter keeps a constant high switching frequency operation not only during heavy load but at light loads. While a circuit of this type improves conduction losses, the circuit still suffers from high switching losses and low light load efficiency, since the circuit operates at the same frequency at both heavy loads and light loads. 
     The shortcomings of the circuit described with respect to  FIG. 3  are overcome in the high light load efficiency synchronous buck generator having pulse skipping control illustrated in  FIG. 5 . The buck regulator  502  includes a voltage source  504 . High side switch  506  comprises a MOSFET transistor having its drain-source path between the voltage source  504  and node  508 . The low side transistor also consists of a MOSFET transistor having its drain-source path connected between node  508  and ground. Each of the high side switch  506  and low side switch  510  have a driver  512 ,  514  connected to their gates and to the UG and LG outputs of the skip mode controller  516 . 
     An inductor  518  is connected between node  508  and the positive input of current-sense amplifier  520 . A resistor  522  is placed across the positive and negative inputs of the current-sense amplifier  520 . The negative input of the current-sense amplifier  120  is connected to the voltage output node  524 . A load resistor  526  is connected in parallel with a load capacitor  528  between voltage output node  524  and ground. Resistor  530  represents a parasitic resistance associated with capacitor  528 . 
     An integrated voltage signal VCOMP is provided by an error amplifier  532  having its negative input connected to the voltage output node  524  and its positive input connected to a reference voltage V REF . The output of the error amplifier  532  is connected to the positive input of a PWM comparator  534  and to ground through a resistor  536  and capacitor  538 . The PWM comparator  534  compares the VCOMP signal from the error amplifier  532  with the sum of the amplified current-sense signal (CSOUT) from the current signal amplifier  520  and a slope compensation signal  536 . The CSOUT signal is also provided as an input to the skip mode controller  516 . The output (COMPOUT) of the PWM comparator  534  is input to the skip mode controller  516 . 
     Referring now to  FIG. 6 , there is illustrated the skip mode controller  516  of the present invention. The skip mode controller  516  receives the CSOUT signal from the current-sense amplifier  520  as one input signal and the COMPOUT signal from the PWM comparator  534  as another input signal. The CSOUT signal is provided to the negative inputs of a first comparator  602  and a second comparator  604 . Comparator  602  detects if the converter  502  is operating in a discontinuous current mode (DCM) or a continuous current mode (CCM) of operation. This is done by comparing the CSOUT input with the zero voltage level signal (VZERO) applied to the positive input of comparator  602 . The output of comparator  602  is applied to an inverter  606 , and the output of invertor  606  is applied to one input of an AND gate  608 . The output of the AND gate  608  comprises output LG which is used to turn on and turn off low side transistor  510 . 
     Comparator  604  determines the pulse skipping current limit threshold for the regulator  502 . This is accomplished by comparing CSOUT with the voltage signal VSKIP which is connected to the positive input of comparator  604 . The output of comparator  604  is connected to one input of an AND gate  610 . The output of AND gate  610  is connected to one input of OR gate  612  having an output connected to the R input of RS flip-flop  614 . The other input of OR gate  612  is connected to the COMPOUT signal from the PWM converter  534 . The COMPOUT signal is also applied to an invertor  616  and to one input of a NAND gate  618 . 
     The output of invertor  616  is provided to the R input of D flip-flop  620 . The D input of D flip-flop  620  is connected to a 5V reference voltage signal, and the CP input is connected to a clock signal. The clock signal is additionally connected to one input of NAND gate  618 . The Q output of D flip-flop  620  is also connected to an input of NAND gate  618 . The output of NAND gate  618  is connected to the S input of RS flip-flop  614 . The Q output of RS flip-flop  614  is connected to the other input of AND gate  608 . The Q output of RS flip-flop  614  provides output signal UG for turning on and off high side transistor  506 . 
     Signals LG and UG are also provided to the inputs of a NOR gate  622 . The output of NOR gate  622  connects to a resistor  624  connected to the CP input of D flip-flop  626 . A capacitor  628  connects between the CP input of D flip-flop  626  and ground. The D input of the D flip-flop  626  is connected to a 5V reference voltage. The R input of D flip-flop  626  connects to LG output signal from AND gate  608 , and the Q output of D flip-flop is connected to the other input of AND gate  610 . 
     The skip mode controller  516  in addition to minimizing conduction losses as will be described in one moment, reduces the switching frequency at light loads and thus the switching losses within the synchronous buck regulator  502 . Within the pulse skipping circuit  516 , the comparator  602  detects if the regulator  502  is operating in either a discontinuous current mode or continuous current mode. Additionally, comparator  604  is used to determine the pulse skipping current threshold for the regulator  502 . The high side switch  506  is turned on when a clock signal pulse is received if the sum of the amplified current signal (CSOUT) and the slope compensation signal  536  is lower than the compensation signal VCOMP from the output of the error amplifier  532 . This process is more fully illustrated in  FIG. 7 . 
     When the high side switch  506  is off at step  702 , inquiry step  704  determines if a clock signal pulse has been received. If not, the high side switch  506  remains off at step  702 . If inquiry step  704  detects a clock pulse, inquiry step  706  determines if the sum of the amplified current-sense signal from the current-sense amplifier  520  and the slope compensation signal  536  are lower than the compensation signal VCOMP from the error amplifier  532 . If not, the high side switch remains turned off. However, if the sum is less than the VCOMP signal, the high side switch  506  is turned on at step  708 . If inquiry step  706  determines that the sum of the amplified current-sense signal and the voltage compensation signal is higher than the compensation signal VCOMP, there is no high side switch  506  on time pulse until the next clock cycle is received. During this time period when no high side switch pulse is provided, the output capacitor  528  provides the load current during this pulse skipping period. 
     Once the high side switch  506  has been turned on, there are two criteria for determining whether the high side switch  506  must be turned off as illustrated in  FIG. 8 . The high side switch  506  is initially on at  802 . Inquiry step  804  determines whether the converter  502  is operating in the DCM or CCM mode. If inquiry step  804  determines that the converter  502  is operating in the DCM mode, inquiry step  806  uses the pulse skipping current (VSKIP/R CS ) to determine when to turn off the high side switch at step  810 . If the regulator gets into DCM, the output of D flip-flop  626  toggles to logic high so that the Rbar input of D flip-flop  614  cannot toggle to logic low until CSOUT is higher than VSKIP. If inquiry step  804  determines that the high side transistor  506  is operating in the CCM mode, inquiry step  808  uses the value of COMPOUT to determine when to turn off the high side transistor at step  810 . If the converter stays at CCM, the output of D flip-flop  626  stays at logic low so that the Rbar input of D-flip flop ( 614 ) is only determined by COMPOUT. 
     Referring now to  FIG. 9 , there are illustrated the switching waveforms associated with the buck regulator  502  and skip mode controller  516 . At time period T 1 , a clock pulse  902  is applied to the regulator  502  and signal UG goes high since the sum of the amplified current signal CSOUT and the slope compensation signal is lower than VCOMP and COMPOUT is high. This turns on the high side switch  506 , and the inductor current I L  begins increasing between T 1  and T 2 . 
     The high side switch  506  is turned off when the inductor current I L  reaches the skipping current limit threshold (VSKIP)  904  because the high side switch  506  is operating in DCM mode according to the output from D-type flip-flop F 1   626 . The high side switch  506  is turned off and low side switch  510  is turned on at T 2  when the inductor current I L  reaches the pulse skipping current limit  904 . 
     When the inductor current I L  is below the zero current limit threshold (V 0 )  906  at T 3 , LG goes low and turns off the low side transistor  510 . This reduces the conduction loss using diode emulation. Both the low side switch  510  and high side switch  506  remain turned off from time period T 3  to T 4 . An additional clock pulse is received at T 4 . However, since the sum of the slope compensation signal and the amplified current-sense signal (CSOUT) are above the loop compensation voltage VCOMP, and COMPOUT remains low, signal UG remains low and the high side and low side switches remain off. This creates a pulse skip while the circuit waits for a next clock cycle. The output capacitor  528  provides the load current during the pulse skipping period  908  between T 4  and T 5 . Use of the pulse skipping period  908  between T 4  and T 5  effectively reduces the switching frequency of the regulator  502  and results in reduced switching losses as well as the reduced conduction losses described above. The reduced switching and conduction losses improves operation of the regulator  502  in light load conditions. 
     Referring now to  FIG. 10 , there is illustrated a comparison of the measured efficiencies between a circuit operating in the pulse skipping mode of the circuit described with respect to  FIG. 6 , and a regulator operating in a forced pulse width PWM mode. As can be seen in  FIG. 10 , a circuit operating in pulse skipping mode can achieve over 80% efficiency at a 10 milliamp load current. A circuit using the forced PWM mode may only achieve 20% efficiency under this same load current. Thus, the pulse skipping PWM control scheme described with respect to  FIGS. 5–9  can significantly improve the light load power conversion efficiency of a voltage regulator thus extending the battery life in portable power applications. 
     Although the preferred embodiment has been described in detail, it should be understood that various changes, substitutions and alterations can be made therein without departing from the spirit and scope of the invention as defined by the appended claims.