Abstract:
A system for boosting the bass of an audio signal to closely match or mirrors a plurality of Robinson-Dadson loudness curves by interpolating coefficients from a table of values representing the Robinson-Dadson loudness. The system having a controller that interpolates the coefficients from the loudness curves and then uses the coefficients in a shelf filter that makes adjustments to the audio signal. The result of the adjustments to the audio signal is the introduction of bass boost slowly through a diminuendo or lowering of level through volume adjustment and to removes the bass boost rapidly during a crescendo or increase in level through user volume adjustment.

Description:
RELATED APPLICATIONS 
     This application claims priority under 35 U.S.C. §119(e) of U.S. Provisional Patent Application No. 60/552,840, filed on Mar. 13, 2004 and titled “SYSTEM AND METHOD FOR VARYING LOW AUDIO FREQUENCIES WITH INTERPOLATED COEFFICIENTS”, and is incorporated by reference in its entirety into this application. 
    
    
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     This application relates generally to audio signals and more practically to boosting the bass content in an audio signal. 
     2. Related Art 
     The results of Fletcher&#39;s and Munson&#39;s research, known as the Fletcher-Munson curves are well known in the art and generally teach that as the level of an audio signal is lowered, the responsiveness of the human ear decreases. The results indicate that at lower volume levels, the human ear is less able to hear the lower frequencies (i.e. bass) in the sound. Presently, many audio systems utilize a manual loudness control to boost low and high-end response at low volume levels to compensate for the responsiveness of the human ear. 
     In  FIG. 1 , an illustration of a set of frequency domain relative level curves  100  commonly referred to as Robinson-Dadson curves is shown. The Robinson-Dadson curves are the result of more recent studies of how the human ear perceives sound and builds upon the original curves developed by Fletcher and Munson in the early 1930&#39;s. These frequency domain relative level curves (equal loudness contours) relate to the frequency response of a human ear to the level of signals being heard. As the signal level decrease, research shows that the responsiveness to the signal by the human ear changes as the bass frequencies decrease. 
     In  FIG. 2 , a set of frequency domain relative level curves  200  illustrates the results from the Robinson-Dadson curves for loudness from 10-90 dB relative to the 90 phon curve. The loudness for 10-90 dB is shown with nine curves at 10 dB, 20 dB, 30 dB, 40 dB, 50 dB, 60 dB, 70 dB, 80 dB, and 90 dB. The 90 dB reference for the Robinson-Dadson curves shows that as the loudness decreases below 90 dB that it is desirable to boost the low frequencies. 
     A known approach to improving the perceived sound quality was proposed in House et al. (U.S. Pat. No. 4,809,338) and implements a bass contour network circuit that is coupled to the program source material. The House et al. patent describes a frequency contour circuit in which the transfer function from source to loudspeaker is altered by a complex attenuation network based on the transfer function of audio reproduction within an automobile. The House et al. patent adds boost to bass frequencies by this approach but the results bare little relationship to Robinson-Dadson curves of  FIG. 2 . In addition, the House et al. patent measures the signal level at the loudspeaker and thus operates in a feedback mode such that adjustments to the signal frequency content affect the measured signal level forming a servo loop. The House et al. patent uses a passive attenuation system that in reality attenuates mid and high frequencies at low volume levels and fails to describe how to restore that lost signal level and uses an average signal level. Other variants on this scheme utilize notch filters for equalizing the frequency resonance within a bounded area, such as a vehicle&#39;s interior. These other variants also use a feedback circuit to detect and adjust bass levels. 
     In another approach, proposed in the Short et al. patents (U.S. Pat. Nos. 4,739,514 and 5,361,381) circuits are implemented that provide automatic loudness compensation to boost the signal in a bandpass centered at 60 Hz through a circuit that utilizes a 2:1 compressor so that input signals can be compressed, filtered, then re-summed into the forward signal path. Similarly, the Werrbach patent (U.S. Pat. No. 5,359,665) describes a low pass filtered signal applied to a compressor and re-summed into the main signal path. Hence both the Short et al. patent and the Werrbach patent responds only to the signal level in the filtered signal path not the full range signal level. 
     In the Kimura patent (U.S. Pat. No. 5,172,358), a multiple pass band control scheme is used. In that scheme, the frequency bands are individually processed. Each frequency band is filtered and the level within the frequency band is detected. The detected level within the frequency band is then used to control the boost level applied to that frequency band using a variable boost limited to that frequency band. Contrary to the Fletcher-Munson curves and the Robinson-Dadson curves, the Kimura patent treats loudness as a concept that applies not to the full audible frequency band of the reproduced signal but to sub-bands at both high and low frequencies. 
     The Iwamura patent (U.S. Pat. No. 5,172,417) describes a three band equalizer that is computed and applied based on reproduced acoustic signal level and applies individual band equalization sections in fixed increments. The Iwamura patent also uses a feedback scheme in which the equalization applied is included in the measured signal that creates a servo-loop in which the compensation chases itself. Further, all these approaches only attempt to simulate the general trend of the Robinson-Dadson curves of  FIG. 1  and  FIG. 2 . 
     These circuits and other known circuits do not mimic the Robinson-Dadson curves and therefore are not accurately responsive to what a listener can hear. Accordingly, there is a need for a circuit that automatically compensates for the decrease in perceived sound levels at lower volumes by mimicking the Robinson-Dadson curves. 
     SUMMARY 
     The system introduces bass boost slowly through a diminuendo or lowering of level through volume adjustment and to removes the bass boost rapidly during a crescendo or increase in level through user volume adjustment. This is done in such a way so that a listener may not notice the boosting action as the volume level is reduced. The changes in audio signals are achieved so that as the volume level or loudness rise, no damage to the audio equipment occurs. 
     A number of parameters associated with curves, such as the Robinson-Dobson curves and are stored in a memory readable by a controller. Each curve has associated coefficients that may be used to adjust a filter that controls the loudness of the lower frequencies of the audio signal. The Robinson-Dadson curves may be closely approximated or mirrored by interpolation between the parameters of at least two curves stored in memory. The interpolation may be used to derive coefficients that result in the filter being configured so the resulting audio signal closely approximate or mirrors the Robinson-Dadson curve. 
     Other systems, methods, features and advantages of the invention will be or will become apparent to one with skill in the art upon examination of the following figures and detailed description. It is intended that all such additional systems, methods, features and advantages be included within this description, be within the scope of the invention, and be protected by the accompanying claims. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The invention can be better understood with reference to the following figures. The components in the figures are not necessarily to scale, emphasis instead being placed upon illustrating the principles of the invention. Moreover, in the figures, like reference numerals designate corresponding parts throughout the different views. 
         FIG. 1  is a frequency domain relative level diagram illustrating the Robinson-Dadson curves. 
         FIG. 2  is a frequency domain relative level diagram illustrating the Robinson-Dadson curves normalized at 1 KHz and relative to the 90 phon curve. 
         FIG. 3  is a block diagram of audio signal processing. 
         FIG. 4  is a block diagram of the DSP of  FIG. 3 . 
         FIG. 5  is a block diagram of the R.M.S. detector of  FIG. 4 . 
         FIG. 6  is a block diagram of the attack and release circuit of the control logic block of  FIG. 4 . 
         FIG. 7  is a block diagram of the attack and release block of  FIG. 6 . 
         FIG. 8  is a block diagram of the coefficient generator of  FIG. 4 . 
         FIG. 9  is a flow diagram of audio signal control of  FIG. 4 . 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       FIG. 3  is a block diagram  300  of audio signal processing. The audio signal  302  is received at an analog-to-digital (A/D) converter  304 . The A/D converter  304  converts the analog audio signal  302  into a digital signal that is received by the control logic block  306  at the digital signal processor (DSP)  308 . The DSP  308  is in data communication with a controller  310  that also resides in the control logic block  306 . The DSP  308  may be implemented as a traditional DSP, a microprocessor, application specific integrated circuit, a circuit that functions as a state machine or any combination of the above listed devices. 
     The controller  310  may receive input from a user interface (not shown) that affects the processing of the input audio signal, such as threshold values and ratio parameters. The received input is then passed to the DSP  308  where the parameters are stored and used. In an alternate implementation, the DSP  308  may implement the functionality of the controller  310  and receive inputs directly from the user interface. 
     The DSP  308  modifies the loudness of the low frequency or bass portion of the digital signal in a way that closely matches or mirrors the Robinson-Dadson curves. The DSP  308  interpolating between stored values of the Robinson-Dadson curves accomplishes the mirroring of the Robinson-Dadson curves. The resulting digital signal from the digital signal processor  308  (and hence the control logic block  306 ) is received at a digital-to-analog (D/A) converter  312 . The D/A converter  312  then converts the digital signal back to an output analog signal  314 . Thus, the processing of the audio signal occurs in the digital domain. The A/D converter  304  and the D/A converter  312  may be implemented within the control logic block  306 . 
     In other implementations, the processing of the audio signal may occur in the analog domain with the control signals occurring in the digital domain. In yet other implementations, the parameters of curves stored in the DSP  308  may be Robinson-Dadson curves, Fetcher-Munson curves, or other parameters that model how the human ear perceives sound. The Robinson-Dadson curves are the result of more recent studies of how the human ear perceives sound, but other curves such as Fletcher-Munson curves may be employed. 
       FIG. 4  illustrates a block diagram of the DSP  308  of  FIG. 3 . The audio signal arrives at a high pass filter  404  and the root mean square (R.M.S.) detector  406 . The audio signal is typically an alternating current (AC) voltage that carries the actual encoded signal. The output of the R.M.S. detector  406  may be in signal communication with the control logic block  408 . The output of the control logic block  408  is shown in signal communication with a coefficient generator  414 . The output of the high pass filter  404  is in signal communication with a shelf filter  416 . The shelf filter  416  also receives coefficients from the coefficient generator  414  and outputs the processed audio signal. 
     The high pass filter  404  filters the audio signal and removes the frequencies below the frequency cut off of the high pass filter  404  from the audio signal. The high pass filter  404  may be a biquad high pass. In other implementations, other types of known high pass filters may be employed. The R.M.S. detector  406  also receives the input audio signal and determines a R.M.S. value that is a measurement of the voltage of the input audio signal. 
     The R.M.S. measurement value of the voltage of the input audio signal may be used as and indication of audio loudness because the R.M.S. value closely indicates the perceived volume level or acoustic power of the input audio signal. The R.M.S. detector  406  produces a direct current (DC) output voltage that is proportional to the R.M.S. level of the input audio signal&#39;s AC voltage. 
     The DC output voltage produced by the R.M.S. detector  406  is passed to the control logic block  408 . The control logic block  408  processes the DC output voltage and converts it into a control parameter that is used to access the coefficient generator  414 . The DC output voltage may be mapped to a digital value. Further, the control logic block  408  maintains the rate of application of boost (i.e. attack time) at a slower rate as relative to the release time (i.e. removal of boost). The threshold values  410  for applying the boost may be set by the user interface and stored in the control logic block  408 . Similarly, the amount  412  or rate of boost may also be set by the user interface and stored in the control logic block  408 . 
     The coefficients generated from the control parameter by the coefficient generator  414  are provided to the shelf filter  416 . The coefficients may be generated by interpolating between the control parameters that are pluralities of values or coefficients that where previously stored or programmed into the memory. The stored pluralities of values or coefficients may represent curves, such as the Robinson-Dadson curves. In another implementation, a set of control parameters associated with a single data set, such as a curve may be stored and other data set derived from the first data set using mathematical equations with interpolation occurring between the two data sets. The shelf filter  416  may be implemented as a biquad shelf filter. The output of the shelf filter  416  may be the output audio signal  314 . 
     Turning to  FIG. 5 , a block diagram  500  of the R.M.S. detector  406  of  FIG. 4  is shown. The R.M.S. detector  406  receives the input audio signal that may have a positive or negative DC voltage value. The absolute value block  502  takes the absolute value of the DC voltage value and determines the magnitude of the voltage of the input audio signal  302 . If the signal has been converted to the digital domain, for example by the A/D  304 , than in an alternate implementation the absolute value block  502  determines the magnitude of the received digital signal. The output of the absolute value block  502  is passed to a low pass filter  504 . 
     The low pass filter  504  acts as an integrator for calculating the R.M.S. level. The logarithm approximation  506  processes the output of the low pass filter  504 . The logarithm approximation  506  enables the signal strengths to be processed in the logarithmic log domain rather than in the linear domain. The R.M.S. output of the logarithm approximation  506  is passed through a scale block  508  and ultimately to the control logic block  408  of  FIG. 4 . The scale block is used to put a lower boundary on the logarithm so that the output of the R.M.S detector  406  has a minimum output. 
     In  FIG. 6 , a block diagram of the attack and release circuit  600  of the control logic block  408  of  FIG. 4  is shown. The R.M.S. output is then received at the control logic block  408 . A comparator  602  that compares a threshold value held in the threshold block  604  to the input from the R.M.S. output by subtracting the threshold value from the R.M.S. value. The threshold value block  604  provides the threshold value that may be set by a user interface via controller  310 ,  FIG. 3 . The threshold value contained in the threshold value block  604  assures that no changes to the low frequency (base) signal occurs if the R.M.S. output is above the threshold value. In other implementations, the threshold value may be hard coded in the threshold value block  604 . 
     A determination is made if the input value is less than zero and if so, it is set to zero in block  606 . The output of block  606  is then adjusted by a ratio set in a ratio block  608 . The ratio is initial set by a user interface via controller  310 ,  FIG. 3 . The ration block  608  may have a hard coded ratio value in other implementations. 
     The adjusted output is then sent to a resistor-capacitor (RC) filter  610  and an attack and release controller  612 . The attack and release controller  612  takes the difference of a control signal that is delayed by the sample delay  614  and the adjusted output. The resulting signal is then used to change the filter coefficients of the RC filter  610 . 
     If the output of the RC filter  610  is greater than the input, then the attack and release controller  612  set the RC Filter  610  to one set of coefficients. If the output is less than the input then attack and release controller  612  set the RC filter  610  to another set of coefficients. This is how the timing of the adding and removing bass boost is controlled. 
     Turning to  FIG. 7  a block diagram of the attack and release controller  612  of  FIG. 6  is shown. The attack and release controller  612  has at least two inputs that may include the adjusted output X(n)  702  and the control signal that is delayed by the sample delay  614  Y(n)  704 . The delayed control signal is subtracted from the adjusted output X(n)  706 . A switch control  708  checks to determine if the difference  706  between X(n)  702  and Y(n)  704  is greater than zero. If the difference  706  is greater than zero in the switch control  708  then an attack condition exist and switch  710  makes a connection with Attack A 1  block  712  that enables the coefficients for the bass boost effect to be more slowly applied relative to the bass boost being removed. If the difference is not greater than zero, than a release condition exists and the switch control  708  makes a connection via switch  710  with the “Release A 1 ” block  714  and the coefficients for the bass boost effect may result in the bass boost being rapidly removed. The resulting coefficients may be directly available as with A 1   716  and may also be combined by a combiner  718  with a scaling value  720  resulting in a scaled coefficient B 1   722 . The switch  710  is shown as a electro-mechanical switch, but may be implemented by any means that provided for a selection between the attack A 1  block  712  and the release A 1  block  714 , including relays, digital switches, and transistors to name but a few examples. 
       FIG. 8  is a block diagram of the coefficient generator  414  of  FIG. 4 . The coefficient generator  414  receives the control signal from control logic  408 ,  FIG. 4 . The control signal may then be scaled by a scaler  802  and passed to a mask  806 . 
     The mask  806  in a fixed-point implementation has a mask of three ones with the rest of the byte being zero. This configuration of the mask  806  results in the most significant bits of the control word generating a number between zero and seven inclusive. In a floating-point implantation, the exponent of the scaled control signal results in the same outcome (a number between zero and seven inclusive). The output of the mask  806  is then shifted by shift block  808  to format the output of the mask into a lookup signal used to select the coefficients within the coefficient generator  810 . 
     The lookup signal may not have coefficients that are directly accessible. In that case, an interpolation occurs within the coefficient generator  810  by an interpolator in order to derive coefficients. The coefficients in the lookup table of the coefficient generator  810  represent the relative loudness curves of  FIG. 2 . The coefficients that may be generated are associated with the 80 dB, 70 dB, 60 dB, 50 dB, and 40 dB levels. In other implementation, other coefficients may be generated or a different number of coefficients may be generated. Generally, two curves are used to determine the coefficients. Each coefficient is generated by the control signal be scaled  804 , . . . , and  814  and combined  812 , . . . , and  816  with the respective control signals. This scaling and combining may occur for each coefficient generated by the coefficient generator  810 . The coefficients are than used by the shelf filter  416  of  FIG. 4  to generate the output audio signal  314 . 
     In  FIG. 9 , a flow diagram  900  of the control logic of  FIG. 4  is shown. The flow starts  902  with an input audio signal  904 . The audio signal may then be filtered  906  by a high pass filter  404 ,  FIG. 4 , to remove the low frequency (bass) signals from the input audio signal. In other implementations, the high pass filter may not be used. The R.M.S. value of the input audio signal is determined  908  and a device such as a R.M.S. detector  406  may be employed. 
     A determination is made  910  if the input audio signal is above a predetermined threshold. The determination is used to decide if the low frequencies require adjusting. If the magnitude of the input audio signal is not above the threshold  910 , then convert the magnitude into a control signal  912 . The control signal is then used to interpolate coefficients from a lookup table that has values associated with a number of predefined curves  914 . The predefined curves may be Robinson-Dadson loudness curves. The coefficients are then used to modify  916  a shelf filter  416 . The shelf filter  416  in turn modifies the input audio signal by boosting the loudness of the bass and processing is complete  918 . The attack time constant (rate of application of boost) may be slow with respect to the release time constant (rate of removal of boost). 
     If the magnitude of the input audio signal is above the predetermined threshold  910 , then no modification of the input audio signal is needed and processing stops  918 . Even though the processing is shown as stopping  918 , in practice it may be implemented in a feedback loop and be a continuous process as long as and input signal is present. 
     While various embodiments of the application have been described, it will be apparent to those of ordinary skill in the art that many more embodiments and implementations are possible that are within the scope of this invention. Accordingly, the invention is not to be restricted except in light of the attached claims and their equivalents.