Abstract:
A trans-impedance amplifier has a front-end circuit including a transistor and collector resistor for setting the open-loop gain of the feedback circuit. The collector resistor, when connected directly to the power supply, has a secondary function of defining the current through the gain transistor, affecting second-order characteristics. A current source is added between the collector resistor and power supply providing a means by which several outside factors can be mitigated, e.g the current source can take over duties for determining/defining the current for the gain transistor, thereby enabling the choice of collector resistor for setting the open-loop gain separate from the current for the gain transistor.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
       [0001]    The present invention claims no priority. 
       TECHNICAL FIELD 
       [0002]    The present invention relates to a circuit for tuning a trans-impedance amplifier for a photodiode, and in particular to the use of a controllable current source for tuning out external effects, e.g. changes in power supply voltage or temperature, on a trans-impedance amplifier. 
       BACKGROUND OF THE INVENTION 
       [0003]    With reference to  FIG. 1 , a conventional TIA circuit, generally indicated at  1 , converts the current IPD exiting a photodiode  2 , into an output voltage VOUT. The photodiode current IPD, which enters the TIA circuit  1  at an input terminal  3 , includes both a DC component and an AC component. The AC component, which carries the information, must be maintained and sent down an amplification chain  4  to final receiving equipment (not shown), while the DC component should be ignored and if possible eliminated, since many front end unit inputs are not designed to tolerate much more than 10 uA. A feedback circuit, generally indicated at  5 , removes the DC component by means of negative feedback implemented by a feedback amplifier  6 /low pass filter (i.e. Capacitor  7 ) and a bypass transistor  8  combination. The feedback amplifier  6 /low pass filter  7  has gain, and removes the AC component of a voltage feedback signal V FB , leaving only a DC component V FBDC . The capacitor  7  is used to set the low-frequency cutoff that the TIA circuit  1  requires. The bypass transistor  8  takes that DC component V FBDC  of the voltage feedback signal V FB  and generates a DC current I FBDC  in the collector  9 , which by the action of negative feedback equals the incoming DC current I PDDC  from the photodiode  2 . Accordingly, the DC component I PDDC  is removed from the incoming signal IPD and passed to the ground GRND through the emitter  10  of the bypass transistor  8 . 
         [0004]    In practice, the TIA circuit can be mounted on a printed circuit board, which forms part of an opto-electronic device, such as a transceiver. The opto-electronic device has an optical connector for coupling to an optical waveguide, e.g. fiber, and an electrical connector for electrically connecting the device to a host computer system. The opto-electronic device can have control and monitoring circuitry; however, the host computer system can also provide control and monitoring systems and functions. 
         [0005]    A conventional trans-impedance amplifier front end circuit  11 , illustrated in greater detail in  FIG. 2 , includes an AC circuit portion  22  having a first amplifying transistor  23  (Q), a first collector resistor  24  (R c ) generally 200 Ohms, which works against the emitter resistance of the first transistor  23  to set the open loop gain A (generally between 10-20), and a first feedback resistor  25  (R fb ) generally 500 ohms. The equation describing the amplifier II is V out /I PD =A/(1+A/R fb ). When the open loop gain (A) is large enough (e.g.&gt;10), the overall gain is approximated by V OUT =I PD ×R fb . A complimentary (DC) circuit  26 , which includes a second amplifying transistor  28  (Q_dc), a second feedback resistor  29  (Rfb_dc), a DC output voltage (out_dc), and a second collector resistor  30  (Rc_dc), is added to provide a reference voltage to the following amplification stages that tracks process variation and environmental effects, such as temperature and power supply voltage changes. The DC component of the out_dc is substantially the same as the out_ac signal over most environmental conditions; however, the DC component is not the focus of the invention. 
         [0006]    The high-speed performance of the trans-impedance front end circuit  11  is sensitive to changes in power supply voltage vdda  27  because the output voltage out_ac is ground referenced. Assuming that the DC current flowing through feedback resistor R fb    25  is negligible, then the DC component of the output voltage out_ac will be equal to the base voltage V be  of the first transistor  23 , independent of power supply voltage vdda from the power supply  27 . When the power supply voltage (vdda) increases, the current through the first transistor  23  ([vdda-out_ac]/Rc) increases. The same is true for the DC circuit  22 . Changes in the power supply voltage vdda causes the operating conditions of the first and second amplifying transistors  23  and  28  (Q and Q_dc) to vary, which in turn causes undesirable variations in performance. Not only does the open loop gain of the amplifier  11  increase with a larger power supply voltage vdda (smaller emitter resistance of the first and second transistors  23  and  28  due to the increased current), but the high-frequency performance of the first transistor  23  changes with changing bias current. Both of these phenomena are generally seen as second-order effects, but still significant to improving real-world performance. The present invention addresses second-order high-speed performance improvements that can be achieved in a trans-impedance amplifier front-end circuit. 
         [0007]    Typically, the design of a trans-impedance amplifier front end circuit involves a trade-off, i.e. the Rc collector resistor  24  sets the open loop gain, but it also defines the current running through the first transistor  23 . The balance is often very difficult to get correct because of changes in the power supply voltage vdda. Often times, the collector resistor  24  Rc, required for the open loop gain, is small (200 Ohms), which for a power supply voltage on the upper end of it&#39;s allowable range causes a large current e.g. 10 mA, to flow through the first transistor  23  in which 2 mA would be desirable. 
         [0008]    U.S. Pat. No. 6,404,281, issued Jun. 11, 2002 in the name of Kobayashi et al; U.S. Pat. No. 6,504,429, issued Jan. 7, 2003 to Kobayashi et al; and U.S. Pat. No. 6,771,132 issued Aug. 3, 2004 to Denoyer et al disclose improvements to TIA feedback circuits that include minimizing the upper limit of the low frequency cut off frequency; however, none of these references provide an adjustable current source for tuning out external effects on the trans-impedance amplifier 
         [0009]    An object of the present invention is to overcome the shortcomings of the prior art by providing an adjustable current source that tunes out external effects on a trans-impedance amplifier. 
       SUMMARY OF THE INVENTION 
       [0010]    Accordingly, the present invention relates to a trans-impedance amplifier circuit, including a front-end circuit, for converting the variable input current signal into an output voltage; and 
         [0011]    wherein the front-end circuit comprises: 
         [0012]    an amplifier circuit with an open loop gain; and 
         [0013]    a controllable current source device which generates a control current for mitigating unwanted effects. 
     
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0014]    The invention will be described in greater detail with reference to the accompanying drawings which represent preferred embodiments thereof, wherein: 
           [0015]      FIG. 1  illustrates a conventional TIA amplifier circuit with feedback circuit; 
           [0016]      FIG. 2  illustrates a conventional TIA front end circuit; 
           [0017]      FIG. 3  illustrates a TIA front end circuit according to the present invention; 
           [0018]      FIG. 4  illustrates another embodiment of TIA front end circuit according to the present invention; and 
           [0019]      FIG. 5  illustrates another embodiment of TIA front end circuit according to the present invention. 
       
    
    
     DETAILED DESCRIPTION 
       [0020]    With reference to  FIG. 3 , by inserting a controlled current device  31 , including a current mirror  32  (Pmirror) and a current source  33  (I) between the power supply  27  (Vdda) and the first collector resistor  24  of the trans-impedance front end circuit  11 , the effect of power supply voltage variation on the performance of the front end circuit  11  can be minimized. The current mirror  32  comprises a third transistor  36  (Pmirror) and a fourth transistor  37  (Pdiode), which replicates the current, e.g. a portion or a multiple thereof, from the tunable current source  33  and generates a voltage in the collector resistance  24  (Rc). In effect, a virtual power supply node is created at the drain of the third transistor  36  that is more stable with respect to ground than the voltage Vdda from the power supply  27 . If the power supply voltage Vdda increases, the voltage across the third transistor  36  increases by the same amount. The new equation describing the current in the first transistor  23  is I FT =[virtual_supply-out_ac]/Rc. However, the current I FT =Icurrentsource/2, assuming that the current mirror  32  is a 1:1 mirror. The division by 2 comes about because half of the current flows through each of the third and fourth transistors  36  and  37 . Accordingly, the voltage virtual_supply=(Icurrentsource×Rc)/2+out_ac, wherein out_ac is effectively the base voltage V be  of the first transistor  23  and is ground referenced. In the illustrated embodiment of  FIG. 3 , the third and fourth transistors  36  and  37  are each a PFET; however, any similar device that can deliver a tunable, controllable current from the positive supply, negative supply, or ground (if the photodiode can tolerate reduced operating voltage) can be substituted. 
         [0021]    There are other ways of accomplishing the controlled current source  31 , but in accordance with the illustrated embodiment, you need a tunable current source  33  and a mirror  32 , which present the current Icurrentsource to the front end circuit  11 . The fourth transistor  37  Pdiode provides the control voltage that tells the current mirror transistor, i.e. the third transistor  36 , Pmirror what current to deliver, which is fundamental current source construction. In use the fourth transistor  37  is the diode, i.e. the side that the reference current (Icurrentsource) enters, and the third transistor  36  is the mirror. If the third and fourth transistors  36  and  37  are identical, then the voltage across the diode, i.e. the fourth transistor  37 , will generate the exact voltage necessary to control the mirror, i.e. the third transistor  36 , to put out the same amount of current. 
         [0022]    The present invention enables a designer to decouple the design aspects of the Rc collector resistor  24  into two components, i.e. separate the setting of the open loop gain A, from the desired amount of current I FT  running through the first transistor  23 . The 2 mA desired bias current I FT  through the first and second transistors  23  and  28  can be set by the current source  33 , and the open loop gain can be set by choice of the collector resistance Rc  24 , in this case, 200 Ohms. 
         [0023]    The embodiment described above is intended to cancel out unwanted power supply variation; however, a designer might also want to mitigate other second order circuit effects, including, but not limited to, temperature, received signal strength, and data rate, by tuning the characteristics of the current source  33 . The emitter resistance of the first transistor  25  is temperature dependent, but the collector Rc resistor  24  may have no temperature coefficient, or the temperature coefficient of the collector resistor  24  may be in the opposite direction to the temperature coefficient of the transistor emitter resistance. Under these conditions, the open-loop gain of the front end circuit  11  will vary greatly from cold to hot temperatures. In these cases, the designer can construct Icurrentsource out of a mixture of currents that are flat with temperature and currents that are proportional to absolute temperature (PTAT) to arrive at a combination that will provide consistent open-loop gain over the designed temperature range. It is well known in the industry that currents of arbitrary temperature coefficient, positive or negative can be generated by adding and subtracting the right proportions of flat and PTAT currents. Therefore, a designer can tune the temperature performance of the front end circuit  11  to achieve any desired performance. A PTAT current may be appropriate for canceling out the decreasing emitter resistance with temperature. A different temperature profile might be desirable for canceling out the temperature coefficient of the collector Rc resistors  24  and  30 . This can be done internal to a control chip provided with the TIA PCB at the time of fabrication as temperature changes in circuit components are generally well modeled. A designers simulations will show the temperature dependence of the open-loop gain, for instance, which can then be mitigated with the appropriate recipe of flat, PTAT and NTAT currents built on-chip. A temperature monitor can also be provided for sending temperature measurements to an external control processor, which tunes the current source  33  accordingly. The latter method is possible, but more expensive to implement and generally not necessary. 
         [0024]    Alternatively, if the open loop gain A is too high at cold temperatures, and too low at hot temperatures, the current (I) from the current source  33  can be made proportional to absolute temperature (PTAT) so that more current could be supplied at hot and less at the cold condition, to ensure the open loop gain A is maintained substantially constant, e.g. ±5%. As above, a temperature monitor can be provided for sending temperature measurements to a control processor, which tunes the current source  33  accordingly. Alternatively, a temperature profile can be predetermined and saved in the control processor to tuning the current source  33 . 
         [0025]    In an alternate embodiment, shown in  FIG. 4 , a current source  40  can be used to pull current (Isupplemental) from the ac_out and dc_out to the ground where the collector resistors  24  and  30  are directly connected to the power supply as in  FIG. 2 . The extra current from the current source superimposes an additional voltage drop across the desired collector resistor  24  Rc (e.g. 200 Ohms). Accordingly, one can get the benefit of a small collector resistor  24  Rc and control over the gain transistor bias current. For a power supply variation of 600 mV and a collector resistance of 200 Ohms, a maximum additional current I supplemental  of 3 mA would be required. The main difficulty with this approach is that a feedback loop is required to sense what the power supply voltage Vdda is so that the correct amount of current can be drawn through the collector resistor Rc  24 . In  FIG. 4 , an opamp  41  is used to sense a voltage that is half of the power supply (vdda/2) and compare it to a voltage reference, in this case 1.5V. A 1.5V reference voltage assumes that the minimum power supply voltage will be 3.0V, so that current will always be drawn through transistors  42  and  43 . The opamp  41  drives an NPN transistor  42 , though any other compatible device can be substituted. If the supply vdda is high, current closer to the maximum 3 mA in the example above is drawn. If the supply vdda is low, Isupplemental closer to 0 mA is drawn. 
         [0026]    In another alternate embodiment similar to the one discussed above with reference to  FIG. 4 , the current Isupplemental is instead pulled by a current source  50  from the pd_anode and inserted between the pd_anode and ground and pulls an appropriate amount of DC bias current through a combination of the feedback resistor  25  (Rfb) and the collector resistor  24  (Rc), i.e. bypassing the first transistor  23  Q. This approach requires less current because the feedback resistor  25  is generally much larger than the collector resistor Rc  24 . The extra current will superimpose the appropriate additional voltage drop on both the collector resistor  24  (Rc) the feedback resistor  25  (Rfb) to make the transistor current in the first transistor  23  consistent with the aforementioned example above. In  FIG. 5 , an opamp  51  is used to sense a voltage that is half of the power supply (vdda/2) and compare it to a voltage reference, in this case 1.5V. A 1.5V reference voltage assumes that the minimum power supply voltage will be 3.0V, so that current will always be drawn through transistors  52  and  53 . The opamp  51  drives an NPN transistor  52 , though any other compatible device can be substituted. As above, if the supply vdda is high, current closer to the maximum 3 mA in the example above is drawn. If the supply vdda is low, a supplemental current (I supplemental ) closer to 0 mA is drawn. 
         [0027]    For a power supply variation of 600 mV, collector resistor  24  of 200 Ohms and feedback resistor  25  of 400 Ohms, the current required would be about 1 mA, ⅓ of the previous embodiment. This implementation suffers one key shortcoming for high-speed applications: the pd_anode node is highly sensitive to capacitance. Each circuit element connected to pd_anode adds capacitance and worsens the performance.