Abstract:
An integrated circuit electrically is supplied with a voltage and includes an output MOS transistor having a gate driven by an output of a logic circuit and a circuit for biasing the gate of the output MOS transistor. The circuit for biasing the gate is provided for lowering a gate-source bias voltage of the output MOS transistor in a conductive state in relation to the gate-source bias voltage that would otherwise be provided by the output of the logic circuit. The present invention is particularly applicable to output stages for I2C buses.

Description:
FIELD OF THE INVENTION 
     The present invention relates to the field of electronic circuits, and, more particularly, to the control of the electrical characteristics of an output transistor in an integrated circuit. 
     BACKGROUND OF THE INVENTION 
     The present invention relates to output transistors, such as MOS transistors, connected to a data bus and presenting a stray capacitance. In particular, the invention relates to controlling a working point in a low state and control time for passing to the low state of such output transistors. 
     To aid in understanding, FIG. 1 shows schematically a conventional output stage  10  of an integrated circuit  20  (implemented with CMOS technology) connected to the input of a data bus  30  of the I2C type. The output stage  10  includes an inverting gate  11 , the output of which drives the gate of an output transistor T OUT . The output transistor T OUT  may be an NMOS transistor, for example. The inverting gate  11  includes a PMOS transistor T 1  and a NMOS transistor T 0  connected by their drains. The source of the PMOS transistor T 1  receives the supply voltage V cc  of the integrated circuit  20 , and the source of the NMOS transistor T 0  is connected to ground (GND). 
     The inverting gate  11  receives as an input (i.e., the gates of the transistors T 0 , T 1  receive) Binary data DT X  to be emitted on bus  30 , and delivers inverted data {overscore (DT)} X  on the gate G of the output transistor T OUT . A bit at 1 at the output of inverting gate  11  corresponds to the application of a voltage V cc  to the gate of the transistor T OUT  (i.e., the transistor T 1  is conductive) and a bit at 0 corresponds to the connection of the gate of transistor T OUT  to ground (i.e., the transistor T 0  is conductive). 
     The transistor T OUT  is connected by its drain D to the data transmission line  31  of the bus  30 , and its source S is connected to ground. The bus  30 , as seen from its input, presents a stray capacitance C BUS  between transmission line  31  and ground. When the transistor T OUT  is in the OFF state, the line  31  is maintained by default at 1 by a bias or pull-up resistor R BUS  receiving a bias voltage V 1 . In practice, the voltage V 1  may be equal to V cc  and is, for example, about 5V. 
     When the transistor T OUT  receives a bit at 1 on its gate (i.e., the voltage V cc ), the transistor T OUT  becomes conductive, the capacitance C BUS  discharges and the output of stage  10  passes to 0. The discharge time is designated T FALL . An output  0  corresponds to a drain-source voltage V DS  and a drain-source current I DS  having stable values designated respectively by V OL  and I OL  and defining the working point in the low state of the transistor T OUT . 
     For historical reasons relating to the use of bipolar transistors in input or output stages of integrated circuits, the industrial specifications of I2C buses require a rather low voltage V OL  on the order of 0.2 to 0.4 V, and a rather high current I OL  on the order of 0.7 to 3 mA. With MOS technology, such a working point may be achieved only with an NMOS transistor having a gate width W much larger than the gate length L, i.e., a ratio W/L clearly larger than 1. 
     With the saturation current I SAT  of a MOS transistor being proportional to the ratio W/L, the transistor T OUT  is thus traversed by a significant current during its switching, the capacitance C BUS  discharges almost instantaneously, and the time for passing to 0 T FALL  is very short. However, the fact that the time T FALL  is very short generates an undesirable electronic noise in transmission line  31 . It is thus desirable that the time T FALL  be lengthened to at least about 20 ns. 
     However, in an output stage  10  as described above, obtaining a rather long time T FALL  is incompatible with obtaining a low voltage V OL  and a rather high current I OL . As a matter of fact, the discharge of the capacitance C BUS  is basically provided by the transistor current in the saturated state, or the current I SAT . Thus, if the ratio W/L of the transistor is decreased to limit the discharge the current I SAT  of the capacitance C BUS , the voltage V OL  is increased at the same time as the current I OL  is decreased. 
     SUMMARY OF THE INVENTION 
     An object of the present invention is to provide a method for lengthening the time T FALL  for passing to zero of an output transistor of the type described above, without modifying its working point V OL , I OL  in the low state. 
     Another object of the present invention is to provide such an output stage including a controller for controlling the fall time of the output transistor when it is connected to a capacitive data bus. 
     These objects are achieved by a method of lengthening the time for passing to zero a signal on a terminal of an output transistor of the MOS type connected to a data transmission line presenting a predetermined electric capacitance while keeping a substantially identical working point in the low state of the transistor and where the gate of the transistor is driven by a logic circuit present in an integrated circuit receiving a predetermined supply voltage. The method includes lowering the gate-source bias voltage of the transistor in the conductive state in relation to the gate-source bias that would otherwise appear at the output of the logic circuit. The transistor may be designed with an increased width over length ratio of the gate of the transistor for keeping the initial working point in the low state. 
     Moreover, the method may include connecting the gate of the transistor to a gate bias circuit. The gate bias circuit may be arranged to reduce the voltage in the high state delivered by the logic circuit on the gate of the transistor. Specifically, the gate bias circuit may include a first MOS transistor substantially identical to the output transistor and arranged as a diode, and a native transistor arranged as a diode and arranged in series with the first MOS transistor. Furthermore, a gate-source bias voltage lower than 2 V may be applied to the output transistor for turning it on. 
     The present invention also relates to an integrated circuit electrically supplied with a predetermined voltage and including an output MOS transistor, a logic circuit providing an output for driving a gate of the output MOS transistor, and a circuit for biasing the gate of output MOS transistor. Accordingly, the gate-source bias voltage of the transistor is lowered in the conductive state in relation to the gate-source bias voltage that would otherwise be present at the output of the logic circuit. 
     More specifically, the gate-source bias voltage of the transistor in the conductive state may be lower than 2 V. The bias circuit may be arranged to reduce the voltage in the high state delivered by the logic circuit. Also, the bias circuit may include at least a first MOS transistor arranged as a diode and substantially identical to the output transistor. The bias circuit may also include a native transistor arranged as a diode and in series with the first MOS transistor. 
     Furthermore, the output transistor may have a width over length ratio of the gate that gives to it a working point in the low state in accordance with the specifications of I2C buses and a fall time of the output signal at least equal to 20 nanoseconds. In practice, the integrated circuit may take the form of an EEPROM memory, for example, and the output stage may be arranged for delivering data read in the memory. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     These objects, characteristics and advantages of the present invention will be described with more detail in the following description of the method according to the invention and an embodiment of an output stage according to the invention, given by way of non-limitative example, in conjunction with the accompanying drawings, in which: 
     FIG. 1, previously described, is a schematic diagram of a conventional series output stage of an integrated circuit according to the prior art; 
     FIG. 2 is a graph illustrating current/voltage curves for a conventional output transistor according to the prior art and an output transistor according to the present invention; 
     FIG. 3 is a schematic diagram illustrating an output stage according to the present invention; 
     FIG. 4 is a schematic diagram of an exemplary embodiment of a bias circuit according to the present invention; and 
     FIG. 5 is a schematic block diagram of an EEPROM memory illustrating an application of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     As noted above, the achievement of a rather long time T FALL  in the output stage  10  represented in FIG. 1 is in practice incompatible with the achievement of a working point including a rather low voltage V OL  and a rather high current I OL . This paradox will be better understood in relation to FIG. 2, which shows the current/voltage graph IDS=f(V DS ) of the output transistor T OUT . When the gate of transistor T OUT  receives voltage V cc  delivered by the transistor T 1  of the inverting gate  11  (FIG.  1 ), the transistor T OUT  progressively turns on. That is, the transistor T OUT  passes from the saturated state to the linear state and its drain-source voltage V DS  slides along graph C 1  from its initial value, which is the bias voltage V 1  of bus  31 , to the value V OL . During the first part of the discharge period of capacitance C BUS , the transistor is in the saturated state because the voltage V DS  is above the differential gate voltage V GS −V T . The current I DS , conventionally designated I SAT , is constant and follows the conventional equation: 
     
       
           I   DS   =I   SAT =( K   P /2)( W/L )( V   GS   −V   T ) 2   (1) 
       
     
     where K P  is a constant (representing mobility of the carriers), V T  is the threshold voltage of the transistor, and V GS  is the gate voltage. The gate voltage V GS  being equal to V cc , equation (1) may also be written: 
     
       
           I   DS   =I   SAT =( K   P /2)( W/L )( V   cc   −V   T ) 2   (2) 
       
     
     During the second part of the discharge period, the voltage V DS  becomes lower than the differential gate voltage V cc −V T . Transistor T OUT  then works in the linear mode, and the drain-source current I DS  follows the conventional equation: 
     
       
           I   DS   =K   P ( W/L )( V   cc   −V   T ) V   DS   (3) 
       
     
     When the capacitance C BUS  is completely discharged, the transistor presents the working point P OL  represented in FIG. 2, defined by the specific values I OL  and V OL  of its drain-source current I DS  and its drain-source voltage V DS . At the working point P OL , equation (3) may be written as: 
     
       
           I   OL   =K   P ( W/L )( V   cc   −V   T ) V   OL   (4) 
       
     
     At the working point P OL , the voltage at the terminals of resistor R BUS  and the voltage at the terminals of transistor T OUT  are bound with the following equation: 
     
       
           V   1   =R   BUS   *I   OL   +V   OL   (5) 
       
     
     Thus, combining equations (4) and (5): 
     
       
           V   1   =R   BUS   I   OL   +I   OL   /[K   P ( W/L )( V   cc   −V   T )]  (6) 
       
     
     It may be deduced therefrom that the value of I OL  is: 
     
       
           I   OL   =V   1   /[R   BUS +1 /[K   P ( W/L )( V   cc   −V   T )]]  (7) 
       
     
     Combining equations (5) and (7), it may also be deduced the value of V OL  is: 
     
       
           V   OL   =V   1 /[1+( R   BUS   K   P ( W/L )( V   cc −V T )]  (8) 
       
     
     In equations (7) and (8), parameters V 1 , V cc , R BUS  are imposed by the environment of the output transistor T OUT . For lengthening the time T FALL , a solution could include decreasing the ratio W/L of the transistor. This would allow the limitation of the discharge current I SAT  of capacitance C BUS  during the saturation period (equation 2), and the limitation of the current I DS  during the linear period (equation 3). However, equations (7) and (8) also show that a decrease of the parameter W/L would lead to a clear increase of voltage V OL  and a small decrease of current I OL . The working point P OL  would thus slide towards a working point P OL ′ on a graph C 2 ′ represented in FIG. 2, which does not satisfy the industrial specifications. 
     Furthermore, the working point P OL  provided by the industrial specifications imposes the ratio W/L of the transistor, and thus imposes the saturation current I SAT . Providing a voltage V OL  and a current I OL  in accordance with the specifications is contradictory with providing a longer discharge time T FALL  which would allow the decrease of the switching noise. 
     The present invention provides for a decrease of the saturation current I SAT  of the output transistor T OUT  without substantially offsetting the working point P OL  Furthermore, the bias voltage VGS of the source of transistor T OUT  is decreased while the ratio W/L is increased (instead of decreasing it). Before describing an example of implementation of this method, the theory of the method will be first explained. 
     The current I OL1  and the saturation current I SAT1  of the conventional transistor T OUT  are given by the following equations (9) and (10) below, in which the characteristics previously indicated by I OL , V OL , W/L, I SAT  are now designated I OL1 , V OL1 , W 1 /L 1 , I SAT1 : 
     
       
           I   OL1   =K   P ( W   1   /L   1 )( V   cc   −V   T ) V   OL1   (9) 
       
     
     
       
           I   SAT1 =( K   P /2)( W   1   /L   1 )(V cc   −V   T ) 2   (10) 
       
     
     Furthermore, the following equations (11) and (12) relate respectively to the current I OL2  and the saturation current ISAT 2  of a transistor according to the invention, having its gate biased by a voltage V 2  different from V cc , and having its drain still biased by the voltage V 1  of bus  30 : 
     
       
           I   OL2   =K   P ( W   2   /L   2 )( V   2   −V   T ) V   OL2   (11) 
       
     
     
       
           I   SAT   2 =( K   P /2)( W   2   /L   2 )( V   2   −V   T ) 2   (12) 
       
     
     According to the invention, it is desired that: 
     
       
           I   OL2   =I   OL1   =I   OL =Constant  (13) 
       
     
     
       
           V   OL2   =V   OL1   =V   OL =Constant  (14) 
       
     
     while satisfying the following equation: 
     
       
           I   SAT2   =N*I   SAT1   (15) 
       
     
     This is because it is desired to lengthen the discharge time T FALL , where N is a number lower than 1 (ranging between 0 and 1). Combining equations (9) (11) (13) and (14), the following equation results: 
     
       
           K   P ( W   1   /L   1 )( V   cc   −V   T ) V   OL   =K   P ( W   2   /L   2 )( V   2   −V   T ) V   OL   (16) 
       
     
     that is: 
     
       
           V   2   −V   T =( W   1   /L   1 )( L   2   /W   2 )( V   cc   −V   T )  (17) 
       
     
     Combining equations (10) (12) and (17): 
     
       
         ( W   2   /L   2 )( V   2   −V   T ) 2   =N ( W   1   /L   1 )( V   cc   −V   T ) 2   (18) 
       
     
     Introducing equation (17) into equation (18): 
     
       
         ( W   2   /L   2 )[( W   1   /L   1 )( L   2   /W   2 )( V   cc   −V   T )] 2   =N ( W   1   /L   1 )( V   cc   −V   T ) 2   (19) 
       
     
     that is: 
     
       
           W   2   /L   2 =(1 /N )( W   1   /L   1 )  (20) 
       
     
     Introducing equation (20) into equation (17) results in the following: 
     
       
           V   2   −V   T   =N ( V   cc   −V   T )  (21) 
       
     
     Thus, according to the invention, decreasing the differential gate voltage V cc −V T  by the factor N to obtain a differential gate voltage V 2 −V T  in response to equation (21), and increasing in parallel, by the inverse factor 1/N, the ratio W 1 /L 1  of a conventional transistor to obtain a ratio W 2 /L 2  in response to equation (20), allows the design of a transistor having a saturation current I SAT2  decreased by factor N in relation to the saturation current I SAT1  of a conventional transistor. This is the case even while keeping the working point P OL  (V OL , I OL ) of the conventional transistor imposed by the industrial specifications. 
     The current/voltage graph C 3  of such a transistor according to the present invention may be seen in FIG.  2 . The graph C 3  presents a saturation current I SAT2  much lower than current I SAT1  (i.e., the flat part of the graph) but keeps a linear part identical to the linear part of graph C 1  when the drain-source voltage V DS  ranges between V OL  and V 2 −V T . Such a transistor limits the saturation current that provides a large part of the discharge of the capacitance C BUS . The increase of the time T FALL  is equal to the ratio between the surface defined by graph C 1  between the voltages V OL , V 1  and the surface defined by graph C 3  between the voltages V OL , V 1 . 
     An output stage  40  according to the present invention is shown in FIG.  3 . The output stage  40  is similar to the stage  10  of FIG. 1 except that a bias circuit according to the invention, represented in the form of a block  41 , is connected between the gate of the transistor T OUT  and ground. When the transistor T 1  of the inverting gate  11  is conductive, the circuit  41  regulates the gate of the transistor and imposes a voltage V 2  in response to equation (21) above. N is chosen to obtain a time T FALL  which does not generate an undesirable switching noise, and T FALL  is preferably above 20 ns. 
     According to an advantageous embodiment illustrated in FIG. 4, the bias circuit  41  includes an element T 3  and a transistor T 4  arranged as diodes and connected in series. The transistor T 4  has substantially the same structure and size as the transistor T OUT . In these conditions, the voltage V 2  at the terminals of circuit  41 , applied to the gate of transistor T OUT , is equal to: 
     
       
           V   2   =V   3   +V   T   (22) 
       
     
     where V T  is the threshold voltage of the diode transistor T 4 , and V 3  is a voltage appearing at the terminals of element T 3 . 
     With this arrangement, the temperature variations of the threshold voltage V T  of transistor T OUT  are partly compensated by the voltage V 2 . The threshold voltage V T  of the transistor T 4  is identical and varies in the same way as the threshold voltage V T  of the transistor T OUT . Thus, the differential gate voltage V 2 −V T  of the transistor T OUT  is given by: 
     
       
           V   2   −V   T =( V   3   +V   T )− V   T   =V   3   (23) 
       
     
     and only depends on the voltage V 3 . The element T 3  may be a reference element providing a stable voltage according to temperature or, more simply, a native NMOS transistor arranged as a diode, the threshold voltage V TNAT  of which is lower that the one of an enhanced transistor. 
     In these conditions, for a voltage V cc  of about 5 V, a voltage V T  of 0.7 V and a voltage V TNAT  of 0.4 V, the voltage V 2  imposed by the bias circuit  41  allows the reduction of about 74% of the differential gate voltage in relation to the value that it would have in the absence of circuit  41 . A decrease in the same proportions of the saturation current I SAT  is also provided. This results from the following calculations: 
     
       
           V   3   =V   TNAT   +V   T   =N ( V   cc   −V   T )  (24) 
       
     
     that is: 
     
       
           N =( V   TNAT   +V   T )/( V   cc   −V   T )  (25) 
       
     
     that is: 
     
       
           N =0.26  (26) 
       
     
     It will be apparent to those skilled in the art that the present invention is likely to have various alternative embodiments and applications. For example, the present invention may also be applied when the bias resistor of the drain of transistor T OUT  is a resistor arranged in the output stage  10  instead of being the resistor R BUS . The drain bias voltage will then be the voltage V cc  instead of the voltage V 1  (although in practice V 1  and V cc  are generally identical). 
     Furthermore, although the bias circuit  41  is shown as connected to the gate of transistor T OUT  in FIG. 3, it is also possible to connect this circuit in another point of the output stage. For example, the inverting gate  11  could be supplied with a regulated voltage lower than voltage V cc  and deliver, in the high state, a gate voltage V GS  in response to equation 21 above. 
     Additionally, although the technical problem having led to the implementation of the present invention relates to buses of the I2C type, the present invention is not limited to such an application and may relate to other data transmission lines of the capacitive type. 
     An application of the present invention to electrically erasable and programmable memories, or EEPROM memories, provided with a serial output for a bus I2C is illustrated in FIG.  5 . The EEPROM memory  50 , represented very schematically in FIG. 5, includes a memory array  51  including memory cells  52  arranged in lines and columns, a line decoder  53 , a column decoder  54 , and a read circuit  55  connected to the column decoder. Read circuit  55  includes sense amplifiers  5 A for delivering binary words W i,j  read in the memory array  51 , e.g., bytes. The binary words W i,j  are applied to a register  56  with a serial output and parallel loading. The serial output of the register  56  is sent to an output terminal  57  of the memory  50  by an output stage  40  as described above.