Abstract:
The invention relates to an IF polyphase filter for filtering received RF signals. The signals are downconverted into intermediate frequency signals before filtering them in the IF polyphase filter. The IF polyphase filter comprises means for defining a passband for the IF polyphase filter. The IF polyphase filter further comprises a passband adapting element for setting the passband of the IF polyphase filter in positive or in negative frequencies. The invention further relates to a receiver comprising the IF polyphase filter according to the invention. The invention further relates to a method for filtering received RF signals by using an IF polyphase filter. The method comprises downconverting the received RF signals into intermediate frequency signals before filtering them in the IF polyphase filter, and defining a passband for the IF polyphase filter. The passband of the IF polyphase filter is set in positive or in negative frequencies.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS  
       [0001]     This application claims priority under 35 USC §119 to Finnish Patent Application No. 20035209 filed on Nov. 14, 2003.  
       FIELD OF THE INVENTION  
       [0002]     The present invention relates to an intermediate frequency (IF) polyphase filter for filtering received radio frequency (RF) signals downconverted into intermediate frequency signals, comprising means for defining a passband for the IF polyphase filter. The invention also relates to a receiver comprising at least an input for receiving RF signals, downconverting means for downconverting the received RF signals into intermediate frequency signals, and an IF polyphase filter for filtering the intermediate signals to separate wanted signals from disturbing signals. The invention also relates to a device comprising a receiver, which comprises at least an input for receiving RF signals, downconverting means for downconverting the received RF signals into intermediate frequency signals, and an IF polyphase filter for filtering the intermediate signals to separate wanted signals from disturbing signals. The invention further relates to a method for filtering received RF signals by using an IF polyphase filter, the method comprising downconverting the received RF signals into intermediate frequency signals before filtering them in the IF polyphase filter, and defining a passband for the IF polyphase filter. The invention also relates to a system comprising a receiver, which comprises at least an input for receiving RF signals, downconverting means for downconverting the received RF signals into intermediate frequency signals, and an IF polyphase filter for filtering the intermediate signals to separate wanted signals from disturbing signals.  
       BACKGROUND OF THE INVENTION  
       [0003]     In some present receivers the bandwidth of the front-end analog IF filter is not calibrated due to minimization of the chip area and the cost.  
         [0004]     This means that typically the IF bandwidth of the receiver is larger than the bandwidth of the actual signal which the receiver is intended to receive (i.e. the wanted signal). When such a receiver is placed in a device comprising a transmitter transmitting signals on a frequency band near the receiving frequency band of the receiver, a disturbing signal (a jamming signal) may exist in the input of the receiver that is outside the actual signal band but still in the received analog band. This is due to the fact that the disturbing signal is not attenuated enough and when the received signals are downconverted by a local oscillator to the IF frequency band also the disturbing signal is downconverted to the IF frequency band. After the downconversion it is almost impossible to separate the disturbing signal from the actual signal.  
         [0005]     At present there are some mobile communication devices which also comprise a satellite positioning system receiver, for example a Global Positioning System (GPS) receiver or a Global Orbiting Navigation Satellite System (GLONASS) receiver. The signal frequencies of the satellite positioning systems are not very far apart from the signal frequencies of, for example, mobile communication systems such as GSM. As a result the transmitter of the mobile communication device may cause disturbing signals to the satellite positioning system receiver. Another cause of jamming can be due to signals generated in the mobile communication receiver. For example, local oscillator signals are generated in the receiver for transforming signals received from a mobile communication network into IF signals. The frequency of the local oscillator signals or some harmonic components of the local oscillator signals or reference crystal oscillator signals may couple to the RF input of the satellite positioning system receiver and can generate spurious signals or other disturbances in the satellite positioning system receiver.  
         [0006]     The above described problem is hard to solve especially in low IF receivers, i.e. receivers in which the IF band is near the baseband. This is due to the fact that the frequency of the local oscillator signal has to be near the frequency of the signals to be received from the satellite positioning system because the difference between the frequency of the signals to be received and the frequency of the local oscillator determine the IF band. Therefore, other strong enough signals laying at a suitable distance from the local oscillator signal can disturb the low IF receiver. Typically the IF band lies around from a few hundred kilohertz up to couple of megahertz and its bandwidth is about 1 to 4 times the bandwidth of the baseband.  
         [0007]      FIG. 1   a  depicts a situation in which a receiver is receiving signals on a certain frequency band i.e. how the receiver “sees” the signals at the front end. These wanted signals are marked with the reference  56  in  FIG. 1   a.  The frequency of the local oscillator (LO) is slightly below the frequency band of the wanted signals and it is marked with the reference numeral  2 . In this example the disturbing signal  55  lies slightly above the frequency band of the wanted signals. When the received signals are downconverted they are shifted to the IF frequency band. This situation is depicted in  FIG. 1   b.  In  FIG. 1   b,  also the pass band of the IF filter is shown and marked with the reference numeral  60 . As can be seen in  FIG. 1   b  the disturbing signal is downconverted inside the pass band of the IF filter. This means that the disturbing signal is also amplified and forwarded to a demodulation stage of the receiver. Thus, the disturbing signal can even hinder the demodulation of the wanted signal or cause distortion to the demodulation result of the wanted signal.  
         [0008]     There are several known ways of implementing a low IF receiver. Firstly, fully real analog signal processing may be used i.e. the signal is treated as a real signal in analog form. This means that a real mixer and real analog bandpass or low-pass filtering are used. The real mixer and real analog bandpass or low-pass filtering operate only with real signals, not with complex signals comprising a real part and an imaginary part. In digital signal processing it is also possible to design the mixers and filters so that they can divide the signal into quadrature components and operate with complex signals. In practice a real bandpass filter is hard or even impossible to realize as an on-chip device for a low IF receiver. Using a real mixer and a real low-pass filter is one solution that yields to a high level of integration but has no image rejection in IF before analog to digital conversion and so leads to stricter requirements for filtering signals in radio frequency band (RF), for example, in the front end stages of the receiver.  
         [0009]     Regarding a receiver, an image frequency is an undesired input frequency that is capable of producing the same intermediate frequency that the desired input frequency produces. The image rejection means that the image frequencies are rejected (or at least significantly attenuated if the full rejection is not possible to achieve).  
         [0010]     Secondly, there is an option to use complex i.e. polyphase analog signal processing. Using a complex mixer and an analog polyphase filter a bandpass function with image rejection can be created. Furthermore, that kind of filter architecture can easily be integrated to an application specific integrated circuit (ASIC) so saving the cost by relaxing the requirements for filtering signals in radio frequency band (RF) and decreasing the number of components outside the ASIC.  
         [0011]     However, one downside of using an on-chip integrated complex mixer and analog polyphase filter compared to an external IF bandpass filter is that, due to process variations, the bandwidth of the filter changes more and so needs to be more oversized, i.e. the bandwidth of the average filter unit needs to be wider than the actual received signal bandwidth and the sharpness of the bandpass of the filter has to be increased in order to provide enough attenuation to signals outside of the bandpass, or calibrated, i.e. the filter has to be tuned to locate the bandpass properly. A disadvantage of the calibration is that structures needed are typically area consuming and in some cases hard to insert into the actual functional design so that the performance is not adversely affected. Also in some signal bands the requirements for the receiver filtering are not so strict meaning that adjacent channel attenuation is not the main parameter that sets the specification.  
         [0012]     For instance, this is the case with GPS signal and calibration of the bandpass function is not necessarily needed but the IF filter band can be oversized so that it meets the specifications regardless of the process variations in the ASIC production. Nevertheless, if the receiver works in a multistandard mobile communication device it needs to be tolerant against possible narrowband interferers.  
         [0013]     A polyphase signal is a vector of independent signals. In this application only a special case of the polyphase signals are considered, namely, two-phase signals. In two-phase system the vectors are two-dimensional and can be represented as follows: 
 
 u ( t )= u   r ( t )+ ju   i ( t ) 
 
 U ( jω )= U   r ( jω )+ jU   i ( jω )   (1) 
 
         [0014]     In Equation (1) u(t) is a two-phase signal in time-domain, u r (t) is the real component of u(t), and u i (t) is the imaginary component of u(t). U(jω) is the signal in the frequency-domain, U r (jω) is the real component of U(jω), and U i (jω) is the imaginary component of U(jω).  
         [0015]     These two-phase signals are also called complex signals. Every frequency component of u(t) can be written as a sum of two sequences. The two sequences of a real signal u r (t) always have the same amplitude and the opposite phase.  
                       A   ⁡     (   ω   )       ⁢     cos   ⁡     [       ω   ⁢           ⁢   t     +     φ   ⁡     (   ω   )         ]         =       ⁢           A   ⁡     (   ω   )       2     ⁢     {       cos   ⁡     [       ω   ⁢           ⁢   t     +     φ   ⁡     (   ω   )         ]       +     j   ⁢           ⁢     sin   ⁡     [       ω   ⁢           ⁢   t     +     φ   ⁡     (   ω   )         ]           }       +                     ⁢         A   ⁡     (   ω   )       2     ⁢     {       cos   ⁡     [       ω   ⁢           ⁢   t     +     φ   ⁡     (   ω   )         ]       -     j   ⁢           ⁢     sin   ⁡     [       ω   ⁢           ⁢   t     +     φ   ⁡     (   ω   )         ]           }                     (   2   )             
 
         [0016]     The first sequence has only a positive frequency component, the second one only a negative frequency component. 
 
 A (ω){ cos [ω t+φ (ω)]+ j  sin [ω t+φ (ω)]}= A (ω) e   jφ(ω)   e   jωt  
 
 A (ω){ cos [ω t+φ (ω)]− j  sin [ω t+φ (ω)]}= A (ω) e   −jφ(ω)   e   −jωt    (3) 
 
         [0017]     The combination of the equations (2) and (3) results in  
                 A   ⁡     (   ω   )       ⁢     cos   ⁡     [       ω   ⁢           ⁢   t     +     φ   ⁡     (   ω   )         ]         =           A   ⁡     (   ω   )       2     ⁢     ⅇ     j   ⁢           ⁢     φ   ⁡     (   ω   )           ⁢     ⅇ     j   ⁢           ⁢   ω   ⁢           ⁢   t         +         A   ⁡     (   ω   )       2     ⁢     ⅇ       -   j     ⁢           ⁢     φ   ⁡     (   ω   )           ⁢     ⅇ       -   j     ⁢           ⁢   ω   ⁢           ⁢   t                   (   4   )             
 
         [0018]     It can be seen from the above that any complex signal A(ω) can be represented as a sum of positive (above 0 Hz) and negative frequency components (below 0 Hz).  
       SUMMARY OF THE INVENTION  
       [0019]     The present invention provides a possibility to configure the passband of a polyphase filter. In detail, it means that the passband of the IF filter can be set to positive or to negative frequencies.  
         [0020]     The invention also provides a complex IF filter based on current summing topology that enables receiving either positive or negative frequency providing image rejection for the unwanted band. In other words, by using the circuits of the present invention it is possible to select the local oscillator of the complex IF receiver working at either a higher or lower frequency than the wanted band.  
         [0021]     According to one aspect of the present invention, there is provided an IF polyphase filter for filtering received RF signals downconverted into intermediate frequency signals, comprising means for defining a passband for the IF polyphase filter. The filter is primarily characterized in that the IF polyphase filter further comprises setting means for setting the passband of the IF polyphase filter in positive or in negative frequencies.  
         [0022]     According to another aspect of the present invention, there is provided a receiver comprising at least an input for receiving RF signals, downconverting means for downconverting the received RF signals into intermediate frequency signals, and an IF polyphase filter for filtering the intermediate signals to separate wanted signals from disturbing signals. The receiver is primarily characterized in that the receiver further comprises setting means for setting the passband of the IF polyphase filter in positive or in negative frequencies.  
         [0023]     According to a third aspect of the present invention, there is provided a device comprising a receiver, which comprises at least an input for receiving RF signals, downconverting means for downconverting the received RF signals into intermediate frequency signals, and an IF polyphase filter for filtering the intermediate signals to separate wanted signals from disturbing signals. The device is primarily characterized in that the device further comprises setting means for setting the passband of the IF polyphase filter in positive or in negative frequencies  
         [0024]     According to a fourth aspect of the present invention, there is provided a method for filtering received RF signals by using an IF polyphase filter, the method comprising downconverting the received RF signals into intermediate frequency signals before filtering them in the IF polyphase filter, and defining a passband for the IF polyphase filter. The method is primarily characterized in that the passband of the IF polyphase filter is set in positive or in negative frequencies.  
         [0025]     According to a fifth aspect of the present invention, there is provided a system comprising a receiver, which comprises at least an input for receiving RF signals, downconverting means for downconverting the received RF signals into intermediate frequency signals, and an IF polyphase filter for filtering the intermediate signals to separate wanted signals from disturbing signals. The system is primarily characterized in that the system further comprises setting means for setting the passband of the IF polyphase filter in positive or in negative frequencies.  
         [0026]     The filter according to a embodiment of the present invention comprises a first and a fourth transconductance amplifier for amplifying the intermediate frequency signals, and a second and a third transconductance amplifier for setting the passband of the IF polyphase filter in positive or in negative frequencies.  
         [0027]     In the filter according to another embodiment of the present invention said setting means comprise means for setting the transconductance of said second and third transconductance amplifier to positive or negative.  
         [0028]     In the filter according to still another embodiment of the present invention said setting means comprise an analog multiplier.  
         [0029]     The present invention has significant advantages compared with prior art solutions. By careful frequency planning the inband interferers can be avoided and the receiver architecture with a controllable intermediate frequency polyphase filter gives some more freedom for the frequency planning of the system by providing a way to get the receiver more tolerant against narrowband interference in e.g. a multistandard environment. A further advantage is that this option can be realized with much simpler control logic and less area than the calibration of the filter would require.  
         [0030]     When compared the filter of the present invention with an external IF filter of the prior art it can be seen that less printed wired board (PWB) area is needed. Also when compared with tuning of the filter the invention achieves savings in on-chip area and, hence, savings in costs. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0031]      FIG. 1   a  shows an example of a frequency spectrum of a wanted signal, a disturbing signal and a local oscillator signal at a front-end of a receiver,  
         [0032]      FIG. 1   b  shows the frequency spectrum of  FIG. 1   a  downconverted to a low IF in the receiver,  
         [0033]      FIG. 1   c  shows another example of a frequency spectrum of a wanted signal, a disturbing signal and a local oscillator signal at a front-end of a receiver,  
         [0034]      FIG. 1   d  shows the frequency spectrum of  FIG. 1   c  downconverted to a low IF in the receiver according to the present invention,  
         [0035]      FIG. 2  is a block diagram of a low IF receiver with configurable polyphase IF filter according to the present invention,  
         [0036]      FIG. 3  shows how the configurable polyphase IF filter may be implemented by using transconductance amplifiers,  
         [0037]      FIGS. 4   a  and  4   b  describe an implementation of transconductance stages according to the present invention using differential pairs,  
         [0038]      FIG. 5  shows an example of an electronic device according to the present invention, and  
         [0039]      FIG. 6  shows as a flow diagram an example of a method according to the present invention. 
     
    
     DETAILED DESCRIPTION OF THE INVENTION  
       [0040]     In the following, the present invention will be described in more detail husing the electronic device  50  of  FIG. 5  as an example. The electronic device  50  comprises a receiver  51  in which a filter  14  according to an embodiment of the present invention is utilized. The details of the receiver  51  and the filter  14  are depicted in  FIGS. 2, 3 ,  4   a  and  4   b.    
         [0041]     Radio frequency signals are received by an antenna  52  and led to the input  1  (the front-end) of the receiver  51  through a bandpass filter  53  ( FIG. 5 ). The bandpass filter  53  is used to filter out signals which are outside the frequency band of the wanted signals. However, the bandwidth of the filter is broader than the bandwidth of the actual signals as was already mentioned above in the description. Referring now to  FIG. 2 , the received signals passed through the antenna coupler  53  are amplified by the low noise high frequency amplifier  10 . After that, the amplified signals are directed to a first input  11 . 1  of a first mixer  11  and to a first input  12 . 1  of a second mixer  12  for mixing the signals with a local oscillator signal  2 . The local oscillator signal  2  is generated by a frequency synthesizer  19  or by another oscillator. In the phase shifter  13  an in-phase local oscillator signal and a quadrature-phase local oscillator signal are generated from the local oscillator signal  2 . The in-phase local oscillator signal is connected to a second input  11 . 2  of the first mixer  11 . The quadrature-phase signal is connected to a second input  12 . 2  of the second mixer  12 . The first mixer  11  performs downconversion of the in-phase signal by mixing the received signal with the in-phase local oscillator signal. At the output  11 . 3  of the first mixer  11  is a downconverted, low IF signal  4  i.e. the I-component of the downconverted signal. The second mixer  12  performs a similar downconversion operation on the quadrature-phase signal by mixing the received signal with the quadrature-phase local oscillator signal. At the output  12 . 3  of the second mixer  12  is a quadrature downconverted, low IF signal  5  i.e. the Q-component of the downconverted signal. The downconverted, low IF signal components  4 ,  5  are fed to an IF filter  14  for filtering the low IF signal components. After the filtering the filtered I-signal component  6  is sampled by a first analog-to-digital converter  15  to form digitized samples of the filtered I-signal component. The filtered Q-signal component  7  is sampled by a second analog-to-digital converter  16  to form digitized samples of the filtered Q-signal component in a similar manner. The I- and Q-samples are then further processed in block  17 . The block  17  represents digital parts of the receiver that are connected to controller and application processor  54  ( FIG. 5 ) through bus  20 . The block  17  comprises, for example, a digital signal processor (DSP) and/or a controller known as such.  
         [0042]     A filter control signal  3  and the frequency of the local oscillator signal  2  must be set in a right manner compared to the wanted channel (frequency band of the signal that is to be received) in order to set the passband of the filter  14  to the wanted channel. The flow diagram  601  of  FIG. 6  discloses some of the steps to control the filter  14 . In the receiver  51  according to the present invention the frequency of the local oscillator signal  2  is set  602  to be either below the wanted channel i.e. RF−IF, or above the wanted channel i.e. RF+IF. The control signal  3  is set to a value with which the passband of the filter  14  is either on negative frequencies, if the frequency of the local oscillator signal is above the wanted channel, or positive frequencies, if the frequency of the local oscillator signal is below the wanted channel. The block  17  determines  603  the correct settings for the passband of the filter  14  and the frequency of the local oscillator signal  2  and uses the filter control signal  3  and the local oscillator control signal  18  for controlling  604 ,  605  the filter  14  and the frequency synthesizer  19 . The signals are then downconverted  606  and filtered  607 .  
         [0043]     Next, the details of an example of the IF filter  14  according to the present invention will be described with reference to  FIGS. 3, 4   a  and  4   b.    
         [0044]     According to an embodiment of the invention illustrated in  FIG. 3  the IF filter  14  is implemented using a transconductance amplifier. For clarity, the figure is presented for single-ended signals but the filter can be realized in differential mode as well. The IF filter  14  has four transconductance amplifier stages  26 - 29 , an in-phase input  21  and a quadrature-phase input  22 , an in-phase output  23  and a quadrature-phase output  24 . The in-phase input  21  is connected to the input of the first transconductance amplifier  26  and the quadrature-phase input  22  is connected to the input of the fourth transconductance amplifier  29 . The output of the first transconductance amplifier  26  is connected to the in-phase output  23  of the IF filter  14  and also to the input of the second transconductance amplifier  27 . Further, the output of the fourth transconductance amplifier  29  is connected to the quadrature-phase output  24  of the IF filter  14  and also to the input of the third transconductance amplifier  28 . There is also a control input  34  in the filter which is connected to a control input of the second transconductance amplifier  27 . The control input signal is also inverted in an inverter  25  to change the sign of the transconductance gm 3  of the third transconductance amplifier  28  opposite to the sign of the transconductance gm 2  of the second transconductance amplifier  27 . The output of the inverter  25  is therefore connected to the control input of the third transconductance amplifier  28 . The absolute value of the transconductance gm 3  of the third transconductance amplifier  28  should be substantially equal to the transconductance gm 2  of the second transconductance amplifier  27 . Further, the transconductance gm 1  of the first transconductance amplifier  26  should be substantially equal to the transconductance gm 4  of the fourth transconductance amplifier  29 .  
         [0045]     The first transconductance amplifier  26  and the fourth transconductance amplifier  29  together with resistors  30  and  32  define the gain of the filter stage. The second transconductance amplifier  27  and the third transconductance amplifier  28  together with resistors  30  and  32  and capacitors  31  and  33  define the center frequency and bandwidth of the filter stage. The control signal from the control input block  34  defines the sign of the transconductance gm 2  of the second transconductance amplifier  27  and the transconductance gm 3  of the third transconductance amplifier  28 . The sign of the transconductance gm 2  of the second  27  and the sign of the transconductance gm 3  of the third transconcuctance amplifier  28  determine whether the passband of the filter stage is located in positive or negative frequencies.  
         [0046]      FIGS. 4   a  and  4   b  present a differential mode implementation of the transconductance amplifiers  26 - 29 . The well-known basic differential pair is drawn in  FIG. 4   a  and this differential pair can be used as the first  26  and the fourth transconductance amplifier  29 . Transistors Q 1  and Q 2  are the actual active elements in the circuit. In this case the transconductance gm 1  of the transconductance amplifier of  FIG. 4   a  is set by the resistors Re 1 , Re 2  and the current source lee 1 .  FIG. 4   b  presents an example of a transconductance amplifier which has a control input for selecting the sign of the transconductance. The sign selection is implemented as an analog multiplier structure. The structure has two switches S 1 , S 2  and an inverter INV. The first switch S 1  is controlled by the filter control signal  3  and the second switch S 2  is controlled by the signal inverted by the inverter INV, i.e. the inverted filter control signal  3 . When the filter control signal  3  has a value which switches the first switch S 1  on, the second switch S 2  is switched off. This structure can be used as the second  27  and the third transconductance amplifier  28  of the filter  14 . Transistors Q 3   p , Q 4   p  and Q 3   n , Q 4   n  form differential pairs that are enabled or disabled by directing the current lee 2  through them by switches S 1  and S 2 . Only one pair at a time is biased i.e. the inverter block INV inverts the value of the select signal wherein only the first switch S 1  or the second switch S 2  is conducting at any given time, depending on the value of the select signal. The transistors Q 3   p  and Q 4   p  with their degeneration resistors form the gm 2  cell for positive frequencies and Q 3   n  and Q 4   n  with their degeneration resistors form the gm 2  cell for negative frequencies. The transconductance gm 2  of the transconductance amplifier of  FIG. 4   b  is defined by the resistors Re 3   p , Re 4   p , Re 3   n , Re 4   n  and the current source lee 2 .  
         [0047]     The filter stage formed by transconductance amplifiers of  FIGS. 4   a  and  4   b  that are connected as shown in  FIG. 3  has the following bandpass function for each of the complex signal branches when the passband is set to positive frequencies:  
                 H   bp     ⁡     (   s   )       =     K     1   -       j   ·     g   m2       ⁢   R     +     s     ω   p                   (   5   )             
 
         [0048]     In the equation (5) H bp (s) means the bandpass transfer function of the filter stage. K is a voltage gain coefficient defined by  26 ,  30 ,  29  and  32  of  FIG. 3  and ω p  is the low-pass equivalent bandwidth set by  30 ,  31 ,  32  and  33  of  FIG. 3 .  
         [0049]     Any lowpass function can be transformed into a complex bandpass function by cascading blocks having transfer function like that described above.  
         [0050]     The passband defined by the pole can be switched to negative frequencies by changing the polarity of the outputs of the second  27  and the third transconductance amplifier  28 .  
         [0051]     In more detail, the voltage transfer function of the filter stage of  FIG. 3  can be expressed as:  
           [             V     out   ,   I       ⁡     (   s   )                   V     out   ,   Q       ⁡     (   s   )             ]     =       [             H   I     ⁡     (   s   )               (       2   ⁢   α     -   1     )     ·       H   Q     ⁡     (   s   )                     (     1   -     2   ⁢   α       )     ·       H   Q     ⁡     (   s   )                 H   I     ⁡     (   s   )             ]     ⁡     [             V     in   ,   I       ⁡     (   s   )                   V     in   ,   Q       ⁡     (   s   )             ]         ,     
     ⁢   where       
       {                     H   I     ⁡     (   s   )       =         g   m1     ·       Z   L     ⁡     (   s   )           1   +       g   m2   2     ·         Z   L     ⁡     (   s   )       2                           H   Q     ⁡     (   s   )       =         g   m1     ·     g   m2     ·       Z   L     ⁡     (   s   )           1   +       g   m2   2     ·         Z   L     ⁡     (   s   )       2                             αε   ⁡     [     0   ,   1     ]                 
 
         [0052]     Vin,I(s) is the voltage at the in-phase input  21  of the filter  14  and Vin,Q(s) is the voltage at the quadrature-phase input  22  of the filter  14 . Vout,I(s) is the voltage at the in-phase output  23  of the filter  14  and Vout,Q(s) is the voltage at the quadrature-phase output  24  of the filter  14 . ZL(s) is the load impedance defined by the resistors  30  and  32  and capacitors  31  and  33 . The binary variable a presents the filter control signal at the control input  34 . By cascading these kind of filter stages any band pass function, e.g. butterworth or chebyshev type, for a complex signal can be realized with the possibility to select positive or negative IF.  
         [0053]     When the receiver  51  is used in a multistandard system jamming signals may exist in the input of the receiver that is outside the actual signal band but still in the received analog band. In a case like that it is useful to have an option of changing the complex IF from positive to negative frequencies or vice versa. The changing can be performed e.g. as follows. It is assumed that there exists a jamming signal which is near and higher than the frequency of the local oscillator signal and, hence, is downconverted to the IF band. If the complex IF is operating on positive frequencies it should therefore be changed to operate on negative frequencies. To achieve this, the frequency of the local oscillator signal  2  is changed to a value which is above the wanted channel and the filter control signal  3  is set to a value which selects the sign of the transconductance gm 2  of the second transconductance amplifier  27  negative and the sign of the transconductance gm 3  of the third transconductance amplifier to a positive value. Respectively, if the complex IF is operating on negative frequencies it should be changed to operate on positive frequencies. To achieve this, the frequency of the local oscillator signal  2  is changed to a value which is below the wanted channel and the filter control signal  3  is set to a value which selects the sign of the transconductance gm 2  of the second transconductance amplifier  27  positive and the sign of the transconductance gm 3  of the third transconductance amplifier to a negative value. The downconverted jamming signal can be moved out of the complex IF filter passband and so it becomes attenuated. The attached  FIGS. 1   a  to  1   d  show in the frequency domain how it happens. In  FIGS. 1   a  and  1   c  the spectrum of the local oscillator signal  2  in the mixer input, the wanted signal  56  and the narrowband jamming signal  55  in the RF input  1  of the receiver  51  are depicted. The difference between  FIGS. 1   a  and  1   c  is that in  FIG. 1   a  the frequency of the local oscillator signal  2  (LO) is below the wanted channel (RF), i.e. the frequency of the local oscillator signal  2  is lower than frequencies of the wanted signals, and in  FIG. 1   c  the frequency of the local oscillator signal  2  (LO) is above the wanted channel (RF), i.e. the frequency of the local oscillator signal  2  is higher than frequencies of the wanted signals. If the local oscillator signal  2  is set to the frequency determined by RF-IF as can be deduced on the basis of  FIGS. 1   a  and  1   b  (LO is below RF and IF is above 0 Hz), and the passband  60  of the IF filter  14  is set to positive frequencies it results a signal spectrum at the IF output  6 ,  7  of the receiver  51  as depicted in  FIG. 1   b  in which the dotted line describes the response of the filter  14 . The jamming signal  55  gets amplified as much as the wanted signal  56 . However, if the frequency of the local oscillator signal  2  is set to RF+IF ( FIG. 1   c ) and the IF filter is set to negative frequencies the situation changes like shown in  FIG. 1   d.  Now the jamming signal  55  gets converted out of the complex IF filter  14  passband and so becomes attenuated compared to the wanted signal  56 .  
         [0054]     The electronic device  50  may also comprise a transmitter  58  and another receiver  57 . The transmitter  58  and the another receiver  57  may be, for example, a transmitter-receiver pair for mobile communication, such as a GSM transmitter-receiver pair. The electronic device of  FIG. 5  also comprises the controller and application processor  54  for controlling the operation of the electronic device, the transmitter  58 , the receivers  51 ,  57 , etc. For example, the controller and application processor  54  instructs the transmitter  58  to transmit signals when necessary. If the transmitter  58  transmits at a frequency channel which may affect that jamming signals are generated at the input  1  of the receiver  51  the controller and application processor  54  informs the block  17  of that. The block  17  then controls the frequency synthesizer  19  to change the frequency of the local oscillator signal  2  and also controls the filter  14  by the filter control signal  3  to change the passband of the filter  14  either to positive or negative frequencies when necessary.  
         [0055]     The electronic device  50  may also comprise means for determining whether external jamming signals exist at the input  1  of the receiver  51 . Such means can comprise, for example, a tunable passband filter (not shown) and a signal strength measuring means (not shown). The signal strength measuring means measure the signal strength at the output of the tunable passband filter. When the passband of the tunable passband filter is near the frequency of the local oscillator signal  2 , the signal strength measuring device indicates if there exists a signal on the passband of the tunable passband filter. The local oscillator may be switched off when the measurement is performed to avoid that the local oscillator signal could be determined as a jamming signal. Another option is that the DSP/control unit  17  of the receiver  51  uses output data of the analog-to-digital converters  15  and  16  to detect the possible jammer. The result of the determination can then be used to decide the necessary changes, if any, to the passband of the filter  14  and to the frequency of the local oscillator signal  2 . In the determination the location of the jamming signal with respect to the wanted signal can be used as the basis for selecting the passband to be either negative or positive and whether the frequency of the local oscillator signal is to be set lower or higher than frequencies of the wanted signals. For example in the situation of  FIG. 1   c  the frequency of the local oscillator signal is higher than frequencies of the wanted signals, near the frequency of the jamming signal. Furthermore, the passband of the filter  14  is (mainly) in negative frequencies. If the jamming signal existed below the wanted signal, the situation would be reversed.  
         [0056]     The electronic device  50  may further comprise a user interface  61  comprising a keypad  61 . 1 , a display  61 . 2  and/or audio means including a codec  61 . 3 , a microphone  61 . 4 , and a speaker  61 . 5 , for example. The electronic device also comprises memory  62 . The electronic device  50  is, for example, a single-mode or a multi-mode mobile communication device with or without a satellite positioning receiver, etc.  
         [0057]     The present invention is not restricted solely to the embodiments presented above, but it can be varied within the scope of the appended claims.