Abstract:
A current control that employs a magnetic amplifier and an active feedback circuit. The feedback circuit establishes the effective operating current of the amplifier at a fixed point. The magnetic amplifier includes a pair of oppositely wound gate windings, a bias winding and a control winding. The gate windings are driven by an oscillator driver that generates a gate winding current and a gate winding voltage. A reference voltage and the gate winding voltage are applied to a feedback amplifier and the feedback circuit. When the gate winding voltage becomes equal to the reference voltage, the feedback circuit is stable and the gate winding current is set to a desired zero current operating point.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     This invention relates generally to a current monitoring and control circuit and, more particularly, to a current monitoring and control circuit that employs a magnetic amplifier and a feedback circuit, where the amplifier circuit provides a bias current that causes the feedback circuit to operate the amplifier at a predetermined point to improve output linearity. 
     2. Discussion of the Related Art 
     Saturable core reactors have been employed in the art as variable impedance devices to detect direct current flowing in an operating circuit, while maintaining isolation between the operating circuit and an output circuit. A saturable reactor is a magnetic circuit element including a coil of wire wound around a magnetic core. The magnetic core significantly alters the behavior of the coil by increasing its magnetic flux and by confining most of the flux to the core. Magnetic flux density (B) is a function of applied magnet motive force (MMF), which is proportional to the ampere turns in the coil. The core includes a plurality of tiny magnetic domains made up of magnetic dipoles. These domains define the magnetic flux that adds to or subtracts from the flux provided by the magnetizing current. After overcoming initial friction, the magnetic domains rotate like small DC motors to become aligned with the applied field. As the MMF is increased, the domains rotate one by one until they all are in alignment and the core is saturated. Eddy currents are induced as the flux changes, causing added loss. 
     Magnetic amplifiers that employ saturable absorbers are known in the art, and are used for various applications, including current monitors for monitoring battery drain in spacecraft telemetry systems. Further, magnetic amplifiers are employed for current control in various systems, such as battery charging circuits and motor control circuits. 
     A conventional magnetic amplifier typically includes two saturable reactors having matched magnetic (permalloy) cores, each being wound with several turns of wires, such as 1500 turns of 38 awg wire. In a magnetic amplifier, each reactor winding is an amplifier gate winding. The two reactor gate windings are coupled in series and have opposing phase, i.e., are wound in opposite directions. The two reactors are positioned side by side and a bias winding, typically about 1000 turns (36 awg or larger), is wound around the reactors. A single control winding extends through the center of the cores, although, multiple turns may be used to increase amplifier sensitivity. The gate windings are coupled in series in opposing phase so that one reactor is reset as the other reactor drives towards saturation. A control current running through the control winding is measured by the magnetic amplifier by magnetic coupling. 
     An alternating current (AC) is applied to the gate windings and the output of the gate windings is full-wave rectified, filtered and resistively loaded to give a DC output voltage proportional to the control current. If the magnetic amplifier operated perfectly, and no current flowed through the control winding, then the current in the gate windings would oppose each other and the DC output voltage would be zero. The control current in the control winding moves the zero axis of the MMF produced by the gate current in the gate windings, and thereby reduces the inductance of the gate windings creating an imbalance between them. The greater the control current in the control winding, the smaller the inductance within a given range of current values. Thus, the greater the control current, the greater the imbalance between the gate windings, and the larger the output voltage. 
     In some applications, the bias winding is shorted or left open, and thus does not affect the magnetic coupling between the control windings and the gate windings. Sometimes it is desirable to shift the zero point of the output voltage when no control current is flowing through the control winding. By applying a bias voltage to the bias winding, the zero point of the output voltage is moved. This has application for determining the direction of the current through the control winding, as will be discussed in more detail below. 
     A magnetic amplifier operates similarly in principle to a current transformer. The ideal current transfer in the amplifier is expressed by:
 
 N   c   I   c   =N   g   I   g   +N   b   I   b 
 
N c  is the number of turns of the control winding, I c  is the control winding current, N g  is the number of turns of the gate windings, I g  is the gate winding current, N b  is the number of turns of the bias winding, and I b  is the bias winding current. The output voltage is V o =I o R o  when it is applied across a fixed load resistor, where:
 
 I   o =( N   c   I   c   −N   b   I   b )/ N   g , and
 
  I   c =( N   g   I   o   +N   b   I   b )/ N   c 
 
       FIG. 1  is a schematic diagram of a conventional magnetic amplifier  10  of the type discussed above. The amplifier  10  includes a control winding  12 , a bias winding  14 , a first gate winding  16  and a second gate winding  18  coupled in series and opposing phase with the gate winding  16 . In this design, the bias winding  14  is shorted and is not used. The amplifier  10  further includes an oscillator driver  24  that drives a transformer  26  with a suitable AC signal. The transformer  26  increases the voltage of the AC signal from the oscillator driver  24 . The secondary winding of the transformer  26  is electrically coupled to the gate windings  16  and  18  and a rectifier  28  including a diode bridge. The AC signal applied to the gate windings  16  and  18  generates the gate winding current I g . The gate winding current I g  is filtered and averaged by a filter  30  including a resistor  32  and a capacitor  34 . Thus, the gate current I g  is rectified, filtered and resistively loaded to provide a DC output voltage V o  representative of the gate winding current I c  that is proportional to the control current I c . 
     When there is no control current I c  in the control winding  12 , the gate winding current I g  is nearly zero because of the equal and opposite windings of the gate windings  16  and  18  are equal and opposite. The control current I c  to be measured is applied to the control winding  12  and alters the gate winding current I g  in the gate windings  16  and  18  by magnetic coupling, as discussed above. Therefore, as the control current I c  increases either in the positive or negative direction, the output voltage V 0  across the resistor  32  increases. 
     Because the gate windings  16  and  18  are driven by a square wave AC signal from the driver  24  and the output voltage V o  is full-wave rectified, the amplifier  10  cannot determine the direction of flow of the control current I c . In other words, a positive or negative control current I c  in the control winding  12  generates the same positive DC output voltage V 0 .  FIG. 2  shows a typical (ideal) control current I c  to output voltage transfer function for a current transducer or magnetic amplifier having an 80 amp operating range. The graph shows the output voltage V 0  in relation to the control current I c  on the control winding  12 , where the control current I c  changes linearly between −80 amps and +80 amps. However, the output voltage V 0  goes from +5 volts to 0 volts, and then back to +5 volts, thus showing that the output voltage V 0  does not identify the polarity of the control current I c . 
     Further, the output voltage V 0  of the amplifier  10  is not linear with respect to the control current I c  applied to the control winding  12 . In other words, changes in the control current I c  are not reflected in changes in the output voltage V 0  in a linear matter. The output linearity is affected by core mismatches and variations in the core construction and gate windings. Also, the effects of magnetizing current non-linearities in the B-H loop winding resistance and winding inductance can introduce errors over the full-scale output. 
     Also, the amplifier  10  is unable to measure a control current I c  on the control winding  12  below the gate winding&#39;s magnetizing current. Particularly, even if the control current in the I c  control winding  12  is zero, leakage in the gate windings  16  and  18  provide a current through the resistor  32  that provides an output voltage V 0 . Therefore, a control current I c  below the magnetizing current of the gate windings  16  and  18  cannot be measured because of system noise. 
     The bias winding  14  responds in a similar manner to the control winding  12  as the gate windings  16  and  18  through magnetic coupling. Because the bias winding  14  has more turns than the control winding  12  (generally 1000:1), a small amount of bias current I b  in the bias winding  14  would produce the same result as a much greater amount of control current I c  in the control winding  12 . The bias winding  14  is generally used to shift the zero current operating point of the amplifier  10  to allow for discrimination of the control current I c  direction. In other words, the output voltage V 0  based on the gate winding current I g  will be some value when a bias applied to the bias winding  14 , but no control current I c  is flowing through the control winding  12 . This discrimination is depicted in  FIG. 3  showing the output relationship of a typical 80 amp magnetic amplifier having a 20 amp offset bias on the bias winding  14 . By applying the bias current I b  to the bias winding  14 , the new zero current operating point of the control winding  12  generates a 1.25 output voltage V 0 . Thus, the bias winding  14  controls the operating point of the amplifier  10 . It is known in the art to provide dual magnetic amplifiers, one including a bias voltage on the bias winding  14 , to provide an indication of the current direction through the control winding  12 . 
     SUMMARY OF THE INVENTION 
     In accordance with the teachings of the present invention, a current control circuit is disclosed that employs a magnetic amplifier and an active feedback circuit. The feedback circuit establishes the effective operating current of the amplifier at a fixed point. The magnetic amplifier includes a pair of oppositely wound gate windings, a bias winding and a control winding. The gate windings are driven by an oscillator driver that generates a gate winding current and a gate winding voltage. A reference voltage and the gate winding voltage are applied to a feedback amplifier in the feedback circuit, and the bias winding is coupled to the source terminal of an FET in the feedback circuit. The output of the feedback amplifier is coupled to the gate terminal of the FET. The drain terminal of the FET provides an output voltage across an output resistor representative of the current flow through the control winding. 
     The output of the feedback amplifier drives the gate terminal of the FET more positive if the gate winding voltage is lower than the reference voltage. When the gate winding voltage becomes equal to the reference voltage, the feedback circuit is stable, and the gate winding current is set to the desired zero current operating point. The bias current is adjusted in response to changes in the current flow through the control winding to maintain the operating point. The changes in the bias current change the voltage across the output resistor that is proportional to the control current in the control winding. The feedback circuit provides linearity between the control current and the gate voltage, and allows the output voltage to provide an indication of the directional polarity of the control current through the control winding. 
     The current control circuit can also employ a reset circuit to insure that the output voltage operates on the correct output voltage slope. Further, the reset circuit can reset the feedback circuit if it goes out of the operating range of the magnetic amplifier. Also, the current control circuit can include output amplifier stages that remove the zero current offset provided by the feedback circuit, provide a signal of the direction of the control current, and provide the desired output range of the control current. 
     Additional advantages and features of the present invention will become apparent from the following description and appended claims, taken in conjunction with the accompanying drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a schematic diagram of a known magnetic amplifier; 
         FIG. 2  is a graph showing current versus voltage transfer function for the magnetic amplifier of  FIG. 1 ; 
         FIG. 3  is a graph showing the output voltage of an 80 amp magnetic amplifier having a 20 amp offset bias; 
         FIG. 4  is a schematic diagram of a current control circuit employing a magnetic amplifier, a feedback circuit, a reset circuit and output amplifier circuits, according to an embodiment of the present invention; and 
         FIG. 5  is a graph showing a feedback circuit output of the current control circuit shown in  FIG. 4  as a function of the control current. 
     
    
    
     DETAILED DESCRIPTION OF THE EMBODIMENTS 
     The following discussion of the embodiments of the invention directed to a current monitoring and control circuit employing a magnetic amplifier and a feedback circuit is merely exemplary in nature, and is in no way intended to limit the invention or its applications or uses. 
       FIG. 4  is a schematic diagram of a current monitoring and control circuit  40 , according to an embodiment of the present invention, that includes a magnetic amplifier  38 . The elements of the magnetic amplifier  38  are the same as the amplifier  10  discussed above and are identified by the same reference numeral. The control circuit  40  includes an active feedback circuit  42  that sets the operating current of the control circuit  40  so that it remains at a fixed operating point defined by a reference voltage V ref , as will be discussed in detail below. By adding the feedback circuit  42 , the full operating range of the circuit  40  is not limited by the range of the amplifier  38 , and can be expanded to the limit of bias circuitry applying a bias voltage V b  to the bias winding  14 . Further, dependencies of the excitation signal quality and external magnetic field effects are virtually eliminated by the feedback circuit  42  so that the output voltage is more linear. Directional sensing (polarity) is inherent in this design because of the offset bias current. 
     The bias winding  14  is coupled to the feedback circuit  42  and receives the bias voltage V b . The bias current I b  through the bias winding  14  is coupled 180° out of phase with the control winding  12 , and therefore acts to cancel the control current I c . The control circuit  40  also includes a negative AC feedback compensation for controlling feedback AC stability. Because of the turns ratio of the bias winding  14  to the control winding  12  (for example, 1000:1), one milliamp of the bias current I b  effectively offsets one amp of the control current I c . In the control circuit  40 , the output voltage of the magnetic amplifier  38  is identified as the gate voltage V g , and the output voltage V 0  is the output of the feedback circuit  42  that is proportional to the control current I c . 
     The feedback circuit  42  includes a feedback comparator or amplifier  44  that receives the reference voltage V ref  at its positive input and the gate voltage V b  across the resistor  32  at its negative input. In one embodiment, the reference voltage V ref  is provided by a precision voltage reference diode  46 , such as an LM 113H diode, and can be, for example, 1.2 volts. The output of the feedback amplifier  44  is coupled to the gate terminal of a field effect transistor (FET)  48 . The source terminal of the FET  48  is coupled to the bias winding  14  and the drain terminal of the FET  48  is coupled to an output resistor  50 , where the output voltage V 0  across the resistor  50  is proportional to the control current I c  in the control winding  12 . Therefore, as the output of the feedback amplifier  44  increases, the gate terminal FET  48  is driven higher, and more of the bias current I b  from the bias winding  14  is allowed to flow through the resistor  50  to generate the output voltage V 0 . 
     In this design, the bias voltage V b  applied to the bias winding  14  is controlled to maintain the gate voltage V g  at the fixed operating point. In other words, as the control current I c  in the control winding  12  changes, the bias voltage V b  is changed so that the gate voltage V b  remains constant at the fixed operating point as set by the reference voltage V ref . The bias voltage V b  is measured across the resistor  50  to determine the control current I c . The bias current I c  cancels the influence of the control current I c  by magnetic coupling in the magnetic amplifier  38 . When the control current I c  is zero, the bias current I b  stabilizes the gate voltage V g  at the desired operating point determined by the reference voltage V ref . If the control current I c  increases in a positive direction, then the bias current I b  is increased to maintain the set point at the output voltage V g , and thus, the output voltage V 0  across the resistor  50  will increase. Likewise, if the control current I c  increases in a negative direction, then the bias current I b  is reduced to maintain the gate voltage V g  at the desired operating point, causing the output voltage V 0  across the resistor  50  to decrease. Therefore, the control circuit  40  can determine the direction of the control current I c  in the control winding  12  because the circuit  40  knows the output voltage V 0  when the control current I c  is zero, and thus, it also knows the direction of the control current I c  by the value of the output voltage V 0  when the control current I c  is not zero. 
     If the gate voltage V g  is at a lower potential than the reference voltage V ref , the gate terminal of the FET  48  is driven more positive. As the gate terminal of the FET  48  is driven more positive, the bias current I b  is increased through the resistor  50 . Because of the gain of the FET  48 , the bias current I b  is drawn through the bias winding  14 . An increase in the bias current I b  shifts the magnetic flux in the gate windings  16  and  18  towards the saturation region, causing an increase in the gate current I g  and an increase in the gate voltage V g . As the gate voltage V g  increases towards the reference voltage V ref , the output of the feedback amplifier  44  goes to zero, reducing the drive power applied to the gate terminal of the FET  48 , and the feedback circuit  42  becomes stable. In the stable mode, the gate voltage V g  is maintained equal to the reference voltage V ref . The bias current I b  is set to the desired zero current operating point of the amplifier  38 . For example, V g  is set to 1.4 volts. 
     When the control current I c  goes more negative, the gate current I g  and the gate voltage V g  tend to decrease. When the gate voltage V g  decreases below the reference voltage V ref , the output of the feedback amplifier  44  drives the gate terminal of the FET  48  more positive. As the gate terminal of the FET  48  is driven more positive, the bias current I b  through the resistor  50  increases. Conversely, when the control current I c  goes more positive, the gate current I g  and the gate voltage V g  tend to increase. When the gate voltage V g  increases above the reference voltage V ref , the output of the amplifier  44  drives the gate terminal of the FET  48  less positive. As the gate terminal of the FET  48  is driven less positive, the bias current I b  through the resistor  50  decreases.  FIG. 5  is a graph showing the output voltage V 0  of the feedback circuit  42  as a function of the control current I c . 
     A large AC ripple could potentially occur on the bias winding  14  at twice its excitation frequency. Therefore, a filtering capacitor  52  is provided to remove this ripple from the output voltage V 0 . Also, a DC bias voltage of I b R b , where R b  is the value of the resistor  50 , must be subtracted from the output voltage V 0  to remove the offset voltage from the output voltage V 0 . 
     As discussed above, the conventional magnetic amplifier cannot differentiate between a positive control current I c  and a negative control current I c . The output voltage V o  will go positive when a negative control current I c  is present and will also go positive when a positive control current I c  is present. The operation of the feedback circuit  42  discussed above allows the control circuit  40  to determine the polarity of the control current I c . However, for the feedback circuit  42  to operate properly, the feedback circuit  42  must always operate on the proper slope of the dual slope ( FIG. 2 ) of the gate voltage V g . A large negative transient control current I c  exceeding the design range of the amplifier  38  can erroneously cause the feedback circuit  42  to try to stabilize on the wrong slope of the gate voltage V g . In other words, if the direction of the control current I c  is changing faster than the response time of the amplifier  38 , the gate voltage V g  may stabilize on the negative slope of the gate voltage V g . This would cause the control circuit  40  to lock up, and not be able to return to the proper operating slope. 
     When the control current I c  is driven more negative by an external load demand, the feedback amplifier  44  and the FET  48  will increase the bias current I b  (out of phase with the control current I c ) to compensate for the increase in the control current and maintain the feedback circuit  42  in the stable condition. When the bias current I b  can no longer increase due to the supply limitations, the useful range of the feedback circuit  42  is exceeded. As the control current I c  continues to increase into the over-range condition, the output voltage V 0  will decrease from the stable reference voltage V ref  to zero volts, and then start to increase more positive towards 1.2 volts on the negative slope of the gate voltage V g . As the gate voltage V g  rises above the reference voltage V ref  causing the output of the feedback amplifier  44  to drive the gate of the FET  48  negative, the feedback circuit  42  will lock into saturation on the wrong (negative) slope of the output voltage V 0 . 
     To protect against this over-range condition, the control circuit  40  includes a reset circuit  54  to detect if the feedback circuit  42  goes out of its operating range, and to return the feedback circuit  42  to its operating range and the proper slope. The reset circuit  54  is necessary in the event the control current I c  changes beyond the rate or amplitude that the feedback circuit  42  can compensate (for example, &gt;2 I o ). If the control circuit  40  is attempting to measure a control current I c  that is out of its operating range, then the reset circuit  54  will repeatedly attempt to reset the feedback circuit  42 , until the control current I c  returns to the operating range of the amplifier  40 . However, if the reset circuit  54  is triggered because the control current I c  is changing its polarity too rapidly, the reset circuit  54  will cause the feedback circuit  42  to return to the proper slope of the gate voltage V g . 
     The reset circuit  54  includes a first comparator  56  and a second comparator  58 . The output of the feedback amplifier  44  is applied to the positive terminal of the first comparator  56 , and the gate voltage V g  is applied to the positive terminal of the second comparator  58 . The reference voltage V ref  is applied to the negative terminals of the comparators  56  and  58 . The output of the first comparator  56  is applied to the negative input of the feedback amplifier  44 , and the output of the second comparator  58  is applied to the negative input of the first comparator  56 . When the output of the feedback amplifier  44  goes below the reference voltage V ref , the output of the comparator  56  goes low. This pulls the negative input of the amplifier  44  low to try and force the output of the feedback amplifier  44  back into the linear control region. The feedback amplifier  44  and the comparator  56  are electrically coupled in a cross-strap configuration so that they continue to toggle until the over-range condition is corrected. 
     When the negative over-range condition is corrected, the positive input of the comparator  58  senses that the gate voltage V g  is reduced below one-half of the reference voltage V ref , and the output of the comparator  58  goes low. The output of the comparator  58  pulls the negative input of the comparator  56  low allowing the output of the comparator  56  to return high, which indicates that it is in the proper operating range of the feedback circuit  42 . The feedback circuit control is reestablished on the correct slope (negative feedback) and the closed loop control operates correctly. Under large transient load conditions, the feedback circuit  42  can transition from the stable slope where negative feedback controls the feedback circuit regulation to the unstable slope where the feedback goes positive. The reset circuit  54  will respond as with the large DC over-current correct condition, correcting the proper slope after the current transient is terminated. An output of the comparator  56  can set a reset flag so that control circuit knows that the reset circuit  54  has been activated. 
     The control circuit  40  also includes an output circuit  62  including a first amplifier stage  64 , a second amplifier stage  66  and a third amplifier stage  70 . The amplifier stages  64 ,  66  and  68  are responsive to the output voltage V 0  and the reference voltage V ref . As will be discussed below, the output circuit  62  removes the zero current offset, identifies the control current I c  polarity, and provides the desired output ranges. 
     The amplifier stage  64  includes an output amplifier  72  that provides an indication of the magnitude of a positive control current I c , such as a positive battery charging current V CHARGE  in the control winding  12 . The positive input of the output amplifier  72  is coupled to the reference voltage V ref  to remove the zero offset bias voltage. The negative input of the output amplifier  72  is coupled to the output voltage V 0 . When the output voltage V 0  is above the reference voltage V ref , the output of the output amplifier  72  is driven to ground indicating the control current I c  is zero or negative. As the control current I c  becomes more positive causing the output voltage V 0  to decrease below the reference voltage V ref , the output of the output amplifier  72  becomes more positive. The output range of the amplifier  72  is set by selecting the value of resistor  74 . In one example, the gain of the amplifier  72  is set for 5 volt full scale output equal to 16 amps of positive charge control current I c . 
     The second amplifier stage  66  includes an output amplifier  76  that an indication of the polarity I DIRECTION  of the control current I c . The positive input of the output amplifier  76  is coupled to the reference voltage V ref , and the negative input of the amplifier  76  is coupled to the voltage output V 0  of the feedback circuit  42 . When the voltage V 0  is greater than the reference voltage V ref , the output of the amplifier  76  is driven low indicating a negative or discharge control current I c . When the control current I c  goes positive, the output voltage V 0  will decrease below the reference voltage V ref , allowing the output of the amplifier  76  to go high to indicate a positive charge current. 
     The third amplifier stage  68  includes a first output amplifier  78  and a second output amplifier  80  that provide an output voltage indication of the magnitude of the negative or discharge control current I c , such as a battery discharge V DISCHARGE  The negative input of the amplifier  78  is coupled to the reference voltage V ref  to remove the zero offset bias voltage, and the positive input of the amplifier  78  is coupled to the output voltage V 0 . As the discharge control current I c  increases, the output voltage V 0  will rise above the reference voltage V ref . Because the amplifier  78  is coupled as a non-inverting stage, the output of the amplifier  78  will increase as its positive input increases with a predetermined gain factor. The output of the amplifier  78  is coupled to the positive input of the amplifier  80 , which is also coupled as a non-inverting amplifier. The output of the amplifier  80  will also increase with a predetermined gain factor as the output voltage V o  increases. 
     In one embodiment, the gain of the amplifiers  72 ,  76 ,  78  and  80  are set to indicate 16 amps of positive charge current and 60 amps of negative charge current. Drifts in the reference voltage V ref  are partially compensated and designed because the voltage V ref  is used to determine the output voltage V 0 , and thus the output current I o , and also to cancel the offset voltage. Thus, with a fixed reference voltage V ref , the resistors  32  and  50  can be used to determine the zero offset current, and the resistor  74  and  82  can be selected to determine the output full-scale current ranges for charge and discharge current. 
     The foregoing discussion discloses and describes merely exemplary embodiments of the present invention. One skilled in the art will readily recognize from such discussion and from the accompanying drawings and claims that various changes, modifications and variations can be made therein without departing from the spirit and scope of the invention as defined in the following claims.