Abstract:
Half-bridge isolation stage topologies are provided in power converters, dividing an input voltage between capacitors. Primary transformer windings are periodically switched across respective capacitors. In current-fed implementations, the current flow through the primary windings is constrained as by an inductive element. In some implementations, a capacitor, primary winding and switch are connected in series in different orders in each of plural legs across the input. Current feed circuitry includes a current constraining component connecting nodes within each of respective legs. The switches of the isolation stage may be turned on with a fixed duty cycle, and a secondary circuit may comprise synchronous rectifiers.

Description:
RELATED APPLICATIONS 
     This application claims the benefit of U.S. Provisional Application Nos. 60/338,658, filed on Nov. 13, 2001, 60/372,621, filed Apr. 12, 2002 and 60/406,272, filed Aug. 27, 2002. The entire teachings of the above applications are incorporated herein by reference. 
    
    
     BACKGROUND OF THE INVENTION 
     Reference is made to U.S. Pat. No. 5,999,417 (the &#39;417 patent), the contents of which are hereby incorporated in this document. 
     The figures of the &#39;417 patent depict power circuit topologies that have isolation stages. Some of these isolation stages are current-fed (for example, through an inductor as shown in FIG. 6A), and others are voltage-fed (for example, from a capacitor as shown in FIG. 6B). In addition, the examples of isolation stages depicted in the &#39;417 patent also have the characteristic that the effective DC voltage seen at the input to an isolation stage (for example, the DC voltage, V B , across capacitor C B  in FIGS. 6A and 6B) is the output voltage multiplied by a transformer&#39;s primary/secondary turns ratio (assuming we ignore resistive drops). Furthermore, because a transformer is operated at about a 50% duty cycle, the primary side switches of the isolation stages depicted in the &#39;417 patent are stressed to approximately 2V B  when they are off. (The actual voltage is slightly higher to insure that the transformer resets before the beginning of the next cycle.) 
     The &#39;417 patent is not limited to these specific isolation stage topologies. Other topologies that exhibit the invention of that patent are also possible. 
     SUMMARY OF THE INVENTION 
     Depending on the application, it may be desirable to replace the isolation stage topologies shown in the &#39;417 patent with their “half-bridge” versions in which the effective DC voltage seen at the input to the isolation stage is twice the output voltage multiplied by the transformer&#39;s turns ratio. In these half-bridge topologies, the primary side switches are stressed to approximately V B , instead of 2V B , and the number of primary turns required on the transformer to achieve a given value of V B  is half the number that is required in the example isolation stage topologies depicted in the &#39;417 patent. 
     Both this lower voltage stress on the switches, and the lower number of primary turns in the transformer can be important. 
     For instance, great improvements have been made in the on-state resistance of MOSFETs that have an off-state voltage rating of 40V and below. If the desired value of V B  is in the 25V to 30V range, the isolation stage topologies of the &#39;417 patent cannot use these low voltage devices, but a half-bridge version of the isolation stage could. 
     Another example of where the half-bridge topology would be advantageous can be found in those DC/DC converters where the output voltage is low, relative to V B , that the number of primary turns required for the isolation stage topologies of the &#39;417 patent is difficult to implement in a transformer of a given size. A half-bridge version of the isolation stage would relieve this problem since only half the number of primary turns would be required. 
     Yet another instance where the lower number of primary windings is useful is in multiple output converters where there are two secondary windings in the transformer, one for each output voltage. Since the ratio of the two secondary windings is determined by the ratio of the two output voltages, these converters often have a very high number of secondary turns, and therefore a very high number of primary turns. If the number of primary windings is too high to implement easily, the factor of two reduction provided by the half-bridge version can be very useful. 
     Therefore, several isolation stage topologies that have the characteristics of a “half-bridge” topology are presented in this document. The various versions include topologies that are either voltage-fed or current-fed, and that have one or two transformers. The concepts and inventions described in the &#39;417 patent apply to these isolation stage topologies in the same manner that they apply to the corresponding isolation stage topologies shown in the &#39;417 patent. However, the present invention is not limited to such applications. 
     The preferred implementations are current-fed implementations in which an input voltage is divided across plural capacitors. Transformer primary windings are periodically switched across the capacitors and current flow through the primary windings is constrained. A secondary circuit is driven from the primary windings to provide an output. 
     A power converter implementing the method has an input which receives an input voltage and an input current from a DC source. Capacitors are connected across the input and divide the input voltage therebetween. Switches periodically switch transformer primary windings across respective capacitors, and current feed circuitry constrains current flow through the primary windings. 
     In certain implementations, two capacitors are connected across the input, each capacitor charging to about ½ the input voltage. Each of the switches conducts about twice the input current. 
     The transformer secondary windings of the secondary circuit may be loosely coupled or on separate transformers; that is, they are not tightly coupled. 
     The current feed circuitry may comprise at least one magnetic element such as an inductor or a transformer. The circuitry may include an inductor which receives the input current as well as another inductor connecting the two capacitors. Current through each primary winding may be constrained by a common component such as an inductor connecting the two windings. A clamp, such as a diode, may be coupled to each switch to limit voltage across the switch when the switch is turned off. 
     The primary circuitry may include a first circuit leg comprising a capacitor, transformer primary winding and switch connected in series across the input. A second circuit leg comprises a capacitor, transformer primary winding and switch connected in series across the input in an opposite order relative to the series connection of the first circuit leg. An inductor interconnects like nodes of the first and second circuit elements. The primary winding may be positioned between the capacitor and switch in each leg. In one embodiment, a current constraining inductor connects nodes between the primary winding and capacitor of each leg, while in another embodiment, the inductor connects nodes between the primary winding and switch of each leg. 
     The switches may be turned on with a fixed duty cycle. The secondary circuit may comprise synchronous rectifiers. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The foregoing and other objects, features and advantages of the invention will be apparent from the following more particular description of preferred embodiments of the invention, as illustrated in the accompanying drawings in which like reference characters refer to the same parts throughout the different views. The drawings are not necessarily to scale, emphasis instead being placed upon illustrating the principles of the invention. 
       FIG.  1 : A Single-Transformer, Voltage-Fed, Half-Bridge Isolation Stage Topology. 
       FIG.  2 : A Two-Transformer, Voltage-Fed, Half-Bridge Isolation Stage Topology. 
       FIG.  3 : A Two-Transformer, Current-Fed, Half-Bridge Isolation Stage Topology. 
       FIG.  4 : Another Two-Transformer, Current-Fed, Half-Bridge Isolation Stage Topology. 
       FIG.  5 : A Single-Transformer, Voltage-Fed, Half-Bridge Isolation Stage Topology. 
       FIG.  6 : A Single-Transformer, Current-Fed, Half-Bridge Isolation Stage Topology. 
       FIG.  7 : Another Single-Transformer, Current-Fed, Half-Bridge Isolation Stage Topology. 
       FIG.  8 : A Single-Transformer, Current-Fed, Half-Bridge Isolation Stage Topology in which Current Flows Through Both Primary Windings during Both Halves of the Cycle. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     A description of preferred embodiments of the invention follows. 
     Half-bridge converters of the voltage-fed type are well known in the prior art.  FIG. 1  shows a half-bridge version of a voltage-fed, single-transformer isolation stage. Two capacitors,  101  and  102 , are placed in series. The midpoint of these two capacitors has a voltage equal to half the overall input voltage, V B . The two switches  103  and  104  alternately connect the transformer&#39;s primary winding  107  across capacitor  101  and capacitor  102 , each for 50% of the cycle. When switch  103  is turned on, the voltage across capacitor  101  (which equals V B /2) is placed positively across the primary winding  107 . When switch  104  is turned on, the voltage across capacitor  102  (which also equals V B /2) is placed negatively across the primary winding. 
     When switch  103  is on and the voltage across the primary winding is positive, controlled rectifier  110  is also on, and current flows to the output through the first secondary winding  108 . In the second half of the cycle when switch  104  is on, controlled rectifier  111  is also on, and current flows to the output through the second secondary winding  109 . 
     At the switch transitions, there is a brief dead time when both switches must be off to avoid the shoot-through that would otherwise occur. Diodes  105  and  106 , which may be the body diodes of switches  103  and  104 , can carry the transformer&#39;s primary current during this dead time. 
     Normally, the two secondary windings in this topology are tightly coupled together. However, they can be made loosely coupled to each other, as indicated with the parasitic inductances  112  and  113  in  FIG. 1 , to achieve the advantages discussed in the &#39;417 patent. A similar setup was shown in the topology of FIG. 9 of the &#39;417 patent since it also used a single transformer. 
     Since the transformer&#39;s primary winding is exposed to only V B /2, instead of V B , the number of primary turns it requires is only half the number it would be for the isolation stage topologies shown in the &#39;417 patent. Similarly, the switches, when they are off, are stressed to approximately V B , instead of approximately 2V B . However, they carry twice the current when they are conducting because the primary winding has half the number of turns. 
       FIG. 2  shows one way to accomplish a half-bridge version of a voltage-fed, two-transformer isolation stage. In this topology, there are two capacitors  201  and  202  in series, each having a DC voltage of V B /2 across it. In parallel with each capacitor is a series combination of a transformer&#39;s primary winding ( 203  or  204 ) and a switch ( 205  or  206 ). Each transformer has a secondary winding ( 209  or  210 ) that is connected to the output capacitor  213  through a synchronous rectifier ( 211  or  212 ). 
     For the first half of the cycle switch  205  is turned on, which connects the primary winding  203  across capacitor  201 . Current flows into primary winding  203 , and out of its corresponding secondary winding  209  to the output. Controlled rectifier  211  is turned on during this time because the secondary winding  210  has a negative voltage across it (as will be explained below) and therefore presents approximately 2V out  to the gate of  211 . 
     Similarly, during the second half of the cycle switch  206  is turned on, which connects the primary winding  204  across capacitor  202 . Current now flows into winding  204  and out of the secondary winding  210  to the output. Controlled rectifier  212  is turned on during this time because the winding  209  has a negative voltage across it. 
     During the half-cycle when switch  205  is off, the magnetizing current of transformer T 1  flows through diode  207 . With this diode turned on, the voltage across the transformer&#39;s primary winding  203  is −V B /2, which permits the transformer to reset for the next cycle. This reset will occur slightly before the next cycle begins since the diode drop and resistive losses make the reverse voltage placed across the winding slightly higher than the forward voltage was. Similarly, transformer T 2  is reset when its magnetizing current flows through diode  208 . 
     At the switch transitions, there is a brief dead time when both switches must be off to avoid the shoot-through that would otherwise occur. Diodes  207  and  208  can carry the transformer&#39;s primary current during this dead time. 
     Since in this topology the primary windings of both transformers T 1  and T 2  are exposed to only V B /2, instead of V B , the number of primary turns they require is only half the number they would be for the isolation stage topologies shown in the &#39;417 patent. Similarly, the switches  205  and  206 , when they are off, are stressed to approximately V B , instead of approximately 2V B . However, they carry twice the current when they are conducting because the primary winding has half the number of turns. 
     Although there are many examples of current-fed, transformer-based converters in the prior art, they do not exhibit the characteristics of a half-bridge topology.  FIG. 3  shows one new way to accomplish a half-bridge version of a current-fed, two-transformer isolation stage. Once again, two capacitors  301  and  302  are placed in series across the input voltage, V B . The DC voltage across each capacitor is V B /2. Switch  305  is connected in series with the primary winding  303  of transformer T 1  and switch  306  is connected in series with the primary winding  304  of transformer T 2 . These two pairs of series elements are then connected to different windings  314  and  315  of transformer T 3 , as shown in the figure. The other ends of T 3 &#39;s two windings are connected to the midpoint formed by capavitors  301  and  302 . 
     The magnetizing inductance of transformer T 3  provides the current-fed feature of this topology. It has a certain DC current flowing through it, just as do the inductors of the current-fed isolation stage topologies shown in the &#39;417 patent. In this case, however, since the “inductor” has two windings, its current can flow through two paths, depending on the state of the isolation stage&#39;s switches. 
     Operation of this circuit is as follows. For the first half of the cycle, switch  305  is turned on and switch  306  is turned off. Transformer T 3 &#39;s magnetizing current flows through winding  314  and through the primary winding  303  of T 1 . This gives rise to a current flowing through the secondary winding  309  of T 1  and the controlled rectifier  311 . For the second half of the cycle, switch  306  is turned on and switch  305  is turned off, and the magnetizing current of T 3  flows through winding  315  and the primary winding  304  of T 2 . Again, this current, reflected by the turns ratio of T 2 , flows through the secondary winding  310  and the controlled rectifier  312 . 
     In both half cycles, the current that flows is dictated by the magnetizing inductance, L M , of T 3 . Both transformers T 1  and T 2  (and their series switches) use the magnetizing inductance of T 3  to provide the current-fed feature. The purpose of the transformer T 3  structure is to allow the inductor controlled current to first flow one way (towards capacitors  301  and  302 ) and then the other way (from capacitors  301  and  302 ), depending on which primary side switch,  305  or  306 , is turned on. 
     During the half cycle that a transformer is not delivering current to the output, it has a negative voltage across it to reset its core. For instance, during the first half cycle when switch  306  is off, the magnetizing current of transformer T 2  flows through diode  308 , which connects the bottom terminal of the primary winding  304  to the voltage V B . The top terminal of primary winding  304  is essentially at the voltage at the midpoint between the capacitors, V B /2. It is different from this only by the small and ac voltage that appears across the winding  315 . The voltage across winding  304  is therefore approximately −V B /2 during the first half of the cycle, which permits transformer T 2  to reset. This reset will occur slightly before the next half cycle begins since the diode drop and resistive losses make the reverse voltage placed across the winding  304  slightly higher than the forward voltage was. Similarly, transformer T 1  is reset when its magnetizing current flows through diode  307 . 
     At the switch transitions, there is a brief overlap time when both switches are on at the same time to ensure that there is a path for the current of T 3  to flow. 
     Since in this topology the primary windings of both transformers T 1  and T 2  are exposed to only V B /2, instead of V B , the number of primary turns they require is only half the number they would be for the isolation stage topologies shown in the &#39;417 patent. Similarly, the switches  305  and  306 , when they are off, are stressed to approximately V B , instead of approximately 2V B . However, they carry twice the current when they are conducting because the primary winding has half the number of turns. 
       FIG. 4  shows another new way to accomplish a half-bridge version of a current-fed, two-transformer isolation stage. In this topology the series combination of capacitor  401 , primary winding  403 , and switch  405  is connected in parallel with the reversed ordered series combination of switch  406 , primary winding  404 , and capacitor  402 . The node where capacitor  201  and primary winding  403  are joined is then connected to the corresponding node where capacitor  202  and primary winding  404  are joined through an inductor,  414 . The total circuit is then placed in series with another inductor  415 . It is these two inductors  414  and  415  that provide the current-fed feature for the topology. 
     For purposes of understanding this topology, note that the DC current “I” flowing through inductor  415  and the DC current flowing through inductor  414  are equal. Also note that both capacitors  401  and  402  have a DC voltage of V B /2 across them, where V B  is the input voltage applied to the entire circuit. 
     The operation of the circuit is as follows. During the first half of the cycle, switch  405  is turned on and switch  406  is turned off. The current flowing through primary winding  403  and switch  405  is approximately 2I (to be exact, we should also acknowledge the ripple currents in the two inductors  414  and  415 ). The voltage across primary winding  403  is approximately V B /2 (the difference is due to the ripple voltage across the capacitors). The current flowing into the primary winding is reflected to the secondary winding  409 , where it flows through the controlled rectifier  411  to the output. 
     During the second half of the cycle, switch  406  is turned on and switch  405  is turned off. Now the current 2I flows through primary winding  404 , and is reflected to the secondary winding  410 , where it flows through the controlled rectifier  412  to the output. The voltage across winding  404  during this half cycle is approximately V B /2. 
     During the half cycle that a transformer is not delivering current to the output, it has a negative voltage across it to reset its core. For instance, during the first half cycle when switch  406  is off, the magnetizing current of transformer T 2  flows through diode  408 , which connects the primary winding  404  across capacitor  402  in a negative manner so that it sees a voltage of −V B /2. This negative voltage permits transformer T 2  to reset. The reset will occur slightly before the next half cycle begins since the diode drop and resistive losses make the reverse voltage placed across the winding  404  slightly higher than the forward voltage was. Similarly, transformer T 1  is reset when its magnetizing current flows through diode  407 . 
     At the switch transitions, there is a brief overlap time when both switches are on at the same time to ensure that there is a path for the currents of inductors  414  and  415  to flow. 
     Since in this topology the primary windings of both transformers T 1  and T 2  are exposed to only V B /2, instead of V B , the number of primary turns they require is only half the number they would be for the isolation stage topologies shown in the &#39;417 patent. Similarly, the switches  405  and  406 , when they are off, are stressed to approximately V B , instead of approximately 2V B . However, they carry twice the current when they are conducting because the primary winding has half the number of turns. 
     Note that the voltage-fed topology of  FIG. 2  can be turned into a single-transformer, voltage-fed topology by combining all four windings on the same core and arranging the polarities of the winding appropriately, as shown in FIG.  5 . 
     Similarly, both of the current-fed topologies of  FIGS. 3 and 4  can be turned into single-transformer, current-fed half-bridge topologies by placing the two primary and two secondary windings all on the same magnetic core and arranging the polarities of the windings appropriately, as shown in  FIGS. 6 and 7 . 
     In all of the topologies shown in  FIGS. 5-7 , current flows into one primary winding, T P1 , and out one secondary winding, T S1 , during the first half cycle. During the second half cycle, the current flows into the other primary winding, T P2 , and out the other secondary winding, T S2 . 
     Alternatively, the current-fed, single-transformer isolation stage of  FIG. 8  could be used. This topology is similar to that shown in  FIG. 7 , except that inductor  814  is connected to different nodes in the circuit of  FIG. 8  than is inductor  714  in the circuit of FIG.  7 . With this alternate connection of the inductor, both primary windings of the single transformer carry current during both half cycles, thereby reducing the conductive losses in the transformer&#39;s primary winding by a factor of two. 
     This is accomplished as follows. During the first half cycle, switch  805  is on and switch  806  is off. The current I flowing through inductor  815  flows through capacitor  801 , primary winding  403  (into the dot), and switch  805 . Similarly, the current I flowing through inductor  814  flows through switch  805 , capacitor  802 , and primary winding  804  (into the dot). A current equal to 2I reflected by the primary to secondary turns ratio then flows through secondary winding  809  (out of the dot) to the output. During the second half cycle when switch  806  is on and switch  805  is off, the two primary windings both carry a current I in the opposite direction (out of the dot), and a current equal to 2I reflected by the turns ratio flows through secondary winding  810  (into the dot) to the output. 
     Note that in all of the single-transformer topologies shown in  FIGS. 5-8 , the two reset diodes (e.g.  407  and  408  in  FIG. 4 ) have been included. Since the single transformer is driven with both a positive and a negative voltage over the course of a full cycle, it is no longer necessary to provide a path for the magnetizing current to flow during a reset half cycle, as it was for the two-transformer topologies. These reset diodes are therefore not needed, although they may still be applied to limit spikes in the voltage across a switch due to a leakage inductance. 
     As was explained in regard to the circuit of FIG. 9 in the &#39;417 patent, the secondary windings of a single-transformer isolation stage would have to be loosely coupled to allow the switch transitions to be made if the control terminals of the controlled rectifiers are driven from signals derived from the secondary windings. Alternatively, the control terminals of the controlled rectifiers in a single-transformer isolation stage could be driven from signals derived from the control circuitry that controls the primary side switches. This control circuitry could be ground referenced to the primary side, or to the secondary side. For either case, the control signal could pass between the two sides of the power circuit through a typical signal-level isolation device  913  such as an opto-isolator or a transformer, as is well known in the art. 
     Once the control signals that synchronize the primary and secondary side switches are passed from one side of the isolation barrier to the other by means other than the main power transformer, it is no longer necessary to ensure that the two secondary windings of the isolation stage are loosely coupled. Therefore, in the single-transformer topologies shown in FIGS.  1  and  5 - 8  of this document (and in FIG. 9 of the &#39;417 patent), the inductors L P1  and L P2 , which represent the leakage and parasitic inductance that keeps the two secondary windings loosely coupled, are no longer needed. The secondary windings can, with this change in how the control signals are synchronized, be tightly coupled. 
     The topologies shown in this document are intended as half-bridge alternatives to the example isolation stage topologies shown in the &#39;417 patent, where the duty cycle is fixed at approximately 50%, and the stage does not provide regulation. However, these topologies can, in general, also be used to provide regulation through the variation of their duty cycle. In particular, the current-fed topologies of  FIGS. 3 and 4  (and their single transformer versions in  FIGS. 6-8 ) can be operated under duty cycle control to regulate the output voltage. If controlled rectifiers are to be used, some other means for driving the control terminals, such as active gate drive circuitry, must be implemented. Or the topology could simply use uncontrolled rectifiers (i.e., diodes). These half-bridge current-fed topologies would then have the characteristics of an isolated up-converter, where the output voltage would equal V B /2 times the turns ratio of the transformer(s) divided by (1-D), where D is the duty cycle. 
     While this invention has been particularly shown and described with references to preferred embodiments thereof, it will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the spirit and scope of the inventions as defined by the appended claims. Those skilled in the art will recognize or be able to ascertain using no more than routine experimentation, many equivalents to the specific embodiments of the invention described specifically herein. Such equivalents are encompassed in the scope of the claims.