Abstract:
A method and circuit for phase and frequency detection having zero static phase error for use in a phase-locked loop system is presented. The phase and frequency detector utilizes a first phase and frequency detector configured to generate first and second pulsed PFD signals. Pulse blocking circuitry is utilized to provide first and second output signals based on the first and second pulsed signals respectively, wherein a time period when both first and second output signals are asserted is substantially reduced from a time period when both first and second pulsed signals are asserted. By reducing the time the first and second output signals are simultaneously asserted, the effects of charge pump current source mismatch are minimized and static phase error is reduced.

Description:
BACKGROUND 
     1. Field of the Invention 
     The present invention relates to phase-locked loops and, more specifically, to phase and frequency detection with little static phase error in phase-locked loop systems. 
     2. Discussion of Related Art 
     Phase-locked loops (“PLLs”) are widely used in modern electronic devices due to their capability of generating an internal feedback clock signal that is phase aligned with an external reference clock signal. PLLs have been utilized in various applications including, for example, cross-chip communications, signal synchronization, data recovery, and frequency modulation. 
     A typical PLL integrates a phase and frequency detector (“PFD”), a charge pump, a low pass filter, and a voltage-controlled oscillator (“VCO”) in a negative feedback closed-loop configuration. The PFD in a PLL receives a reference clock signal and an internal feedback clock signal and generates two pulsed signals based on the detected phase difference between the reference clock and internal feedback clock signal. These pulsed signals drive the charge pump to adjust the control voltage provided to the VCO, thereby changing the frequency of the signal, output by the VCO. In current PFD implementations, the level of both pulsed signals generated by the PFD may be set to a high logic level during a period when no charge should be injected by the charge pump. In such an instance, if the source and sink current sources are perfectly matched, the net charge injected by the charge pump is ideally zero. Actual charge pump current sources, however, often exhibit some mismatch, causing the internal feedback clock signal generated by the VCO to shift in phase from its ideal location. Non-ideal phase shift attributed to mismatched charge pump current sources is called static phase error. Static phase error may be reduced by minimizing the period during which the charge pump source and sink current sources simultaneously inject charge. 
     It is desirable to develop a novel and improved PFD that reduces static phase error and relaxes matching requirements of charge pump currents. 
     SUMMARY 
     In accordance with some embodiments of the present invention, a phase and frequency detector includes a first phase and frequency detector configured to generate first and second pulsed signals in response to a comparison between a defined occurrence of first and second input signals; and a pulse blocker that receives the first and second pulsed signals and provides first and second output signals, wherein a time period when both first and second output signals are asserted is substantially reduced from a time period when both first and second pulsed signals are asserted. 
     In some embodiments the first phase and frequency detector may comprise first and second D-type flip-flops, wherein the clocking terminals of the first and second D-type flip-flops receive the first and second input signals respectively, the D terminals of the first and second D-type flip-flops are set to an asserted state, and the Q outputs of the first and second D-type flip-flops provide the first and second pulsed signals respectively; and a reset signal generator for asserting a reset signal provided to the reset terminals of the first and second D-type flip-flops based on the state of the first and second pulsed signals. Further, in some embodiments, the pulse blocker may comprise first and second NAND gates, wherein the first NAND gate is enabled by the first pulsed signal and the output of the second NAND gate, and the second NAND gate is enabled by the second pulsed signal and the output of the first NAND gate; a first inverter configured to invert the output of the first NAND gate and provide the first output signal; and a second inverter configured to invert the output of the second NAND gate and provide the second output signal. 
     In accordance with some embodiments of the present invention, a method for detecting the phase difference between a first and a second input signal includes generating first and second pulsed signals based on the first and second input signals, the first pulsed signal being switched to a second state from a first state in response to a defined occurrence of the first input signal, the second pulsed signal being switched to the second state from the first state in response to the same defined occurrence of the second input signal, and the first and second pulsed signals being switched from the second state to the first state after a certain delay period following both of the first and second pulsed signals reaching the second state; and generating first and second output signals based on the first and second pulsed signals respectively, such that in response to the first pulsed signal reaching the second state prior to the second pulse signal reaching the second state, the first output signal is switched to the second state from the first state for the period after the first pulsed signal reaches the second state and before the second pulsed signal reaches the second state, and in response to the second pulsed signal reaching the second state prior to the first pulsed signal reaching the second state, the second output signal is switched to the second state from the first state for the period after the second pulsed signal reaches the second state and before the first pulsed signal reaches the second state. 
     Further embodiments and aspects of the invention are discussed with respect to the following figures, which are incorporated in and constitute a part of this specification. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  illustrates a schematic diagram of a charge pump PLL in accordance with some embodiments of the present invention. 
         FIG. 2   a  illustrates a schematic diagram of a PFD in accordance with some embodiments of the present invention. 
         FIG. 2   b  illustrates an exemplary signal timing diagram of the PFD illustrated in  FIG. 2   a  in accordance with some embodiments of the present invention. 
         FIG. 3  illustrates an exemplary signal timing diagram of a PLL that utilizes the PFD shown in  FIG. 2   a  in locked status wherein the PLL charge pump current sources are mismatched, in accordance with some embodiments of the present invention. 
         FIG. 4   a  illustrates a schematic diagram of a PFD that includes pulse blocking circuitry in accordance with some embodiments of the present invention. 
         FIG. 4   b  illustrates an exemplary signal timing diagram of the PFD that includes pulse blocking circuitry illustrated in  FIG. 4   a  in accordance with some embodiments of the present invention. 
         FIG. 5  illustrates an exemplary signal timing diagram a of PLL that utilizes the PFD shown in  FIG. 4   a  in locked status, in accordance with some embodiments of the present invention. 
     
    
    
     In the figures, elements having the same designation have the same or similar functions. 
     DETAILED DESCRIPTION 
       FIG. 1  illustrates a schematic diagram of a charge pump PLL  100  in accordance with some embodiments of the present invention. Charge pump PLL  100  includes PFD  106 , charge pump  112 , low pass filter  124 , and VCO  126  in a negative feedback closed-loop configuration. In some embodiments, charge pump PLL  100  may include frequency divider  130  in the feedback loop path between VCO  126  and PFD  106 . 
     PFD  106  receives reference clock signal  102  and internal feedback clock signal  104 , generating pulsed signals UP  108  and DN  110 . In some embodiments, the relative pulse width of signals UP  108  and DN  110  is proportional to the phase difference between reference clock  102  and internal feedback clock  104  as detected by PFD  106 . Charge pump  112  receives signals UP  108  and DN  110  from PFD  106  and injects charge using source current source  114  and/or sink current source  116  based on the levels of signals UP  108  and DN  110 . For example, when UP  108  is set to a high logic level, charge pump  112  switch  118  may cause source current source  114  to inject charge at PLL node  122 . Similarly, when DN  110  set to a high logic level, charge pump  112  switch  120  may cause sink current source  116  to inject negative charge at node  122 . 
     Charge injected by charge pump  112  into PLL node  122  is filtered by low pass filter  124  and fed to the control input of VCO  126 . VCO  126  generates VCO output signal  128  having a given frequency based on the voltage provided to the control input of VCO  126 . Frequency divider  130  divides the frequency of VCO output signal  128  by an integer N to generate internal feedback clock signal  104 . Internal feedback clock signal  104  is coupled to PFD  106  as an input signal, thereby forming a negative feedback loop. In some embodiments, VCO output signal  128  may be fed directly to PFD  106  as a reference input. 
     During PLL operation, if internal feedback clock signal  104  is not phase aligned with reference clock signal  102 , PFD  106  drives charge pump  112  via signals UP  108  and DN  110  to adjust the voltage provided to the control input  122  of VCO  126 . Charge pump  112  adjusts the voltage provided to the control input of VCO  126  accordingly until internal feedback clock signal  104  is phase aligned with reference clock signal  102  (i.e., phase locked). Once internal feedback clock signal  104  is phase aligned with reference clock signal  102 , VCO output  128  and/or internal feedback clock signal  104  may be used to synchronize system events with reference clock signal  102 . 
       FIG. 2   a  illustrates a schematic diagram of a PFD  200   a  in accordance with some embodiments of the present invention. PFD  200   a  includes two D flip-flops (“DFFs”)  206 ,  208 , AND gate  212 , and delay buffer  216 . The D input of DFF  206  is coupled to signal  202  set to a constant high logic level. Similarly, the D input of DFF  208  is coupled to signal  204  set to a constant high logic level. In some embodiments, a single signal set to a constant high logic level may be coupled to the D inputs of both DFFs  206 ,  208 . The clock input of DFF  206  is coupled to PLL reference clock signal  102 . Similarly, the clock input of DFF  208  is coupled to internal feedback clock signal  104  provided by VCO  126  of PLL system  100  via the feedback loop. 
     PFD  200   a  output UP  108  is provided by the Q output of DFF  206 . Similarly PFD  200   a  output DN  110  is provided by the Q output of DFF  208 . PFD  200   a  output signals UP  108  and DN  110  are coupled to AND gate  212  as inputs. Delay buffer  216  receives the output  214  of AND gate  212  and provides the reset signal  210  for DFFs  206  and  208 . 
       FIG. 2   b  illustrates an exemplary signal timing diagram  200   b  of PFD  200   a  in accordance with some embodiments of the present invention. The operation of PFD  200   a  is described below with reference to signal timing diagram  200   b . For illustrative purposes a situation where reference clock signal  102  leads internal feedback clock signal  104  is described. PFD  200   a , however, operates similarly when internal feedback clock signal  104  leads reference clock signal  102 . As described, DFFs  206  and  208  are configured to capture D inputs  202  and  204  at the rising edges of reference clock  102  and internal feedback clock  104  respectively. However, in some embodiments, DFFs  206  and  208  may be configured to capture according to the falling edges of their respective clock signals. 
     The Q output of DFF  206 , corresponding with PFD  200   a  output signal UP  108 , captures the state of D input signal  202  at every rising edge of reference clock signal  102 . As DFF  206  D input signal  202  is set to a constant high logic level, at every rising edge of reference clock signal  102 , PFD output UP  108  is set to a high logic level after a period corresponding to the inherent capture delay time of DFF  206 . Similarly, the Q output of DFF  208 , corresponding with PFD  200   a  output signal DN  110 , captures the state of D input signal  204  at every rising edge of internal feedback clock signal  104 . As DFF  208  D input signal  204  is set to a constant high logic level, at every rising edge of internal feedback clock signal  104 , PFD output DN  110  is set to a high logic level after a period corresponding to the inherent capture delay time of DFF  208 . The phase mismatch between reference clock signal  102  and internal feedback clock signal  104 , t 1  and t 2 , is shown with respect to their rising signal edges. Assuming that DFF  206  and DFF  208  exhibit the same inherent capture delay time, the phase mismatch between UP  108  and DN  110 , measured with respect to their rising edges will also be t 1  and t 2 . 
     When signals UP  108  and DN  110  are set to high logic levels by DFF  206  and DFF  208  respectively, AND gate  212  output signal  214  is set to a high logic level after a period corresponding to the inherent delay time of AND gate  212 . AND gate output signal  214  is delayed by delay buffer  216  and the delayed AND gate output signal  214  is provided to DFF  206  and DFF  208  as reset signal  210 . Once reset signal  210  is asserted and after the inherent reset delay time of the DFFs  206  and  208 , DFF  206  Q output UP  108  and DFF  208  Q output DN  110  reset to a low logic level. Output UP  108  and DN  110  and remain at this level until DFF  206  and DFF  208  capture the signal levels at inputs  202  and  204  at the next rising clock edges of reference clock signal  102  and internal feedback clock signal  104  respectively. 
     The differential width t 1  between PFD  200   a  output UP  108  and DN  110  is proportional to the phase difference between reference clock signal  102  and internal feedback clock signal  104 . However, in this PFD implementation, output signals UP  108  and DN  110  are asserted simultaneously for a period corresponding to the cumulative delay time of AND gate  212 , delay buffer  216 , and the reset time of DFFs  206  and  208 . Due to current source mismatch in charge pump  112 , as described below with reference to  FIG. 3 , when signals UP  108  and DN  110  are both asserted, charge pump  112  may inject charge when ideally no net charge should be injected. 
       FIG. 3  illustrates an exemplary signal timing diagram  300  of a PLL  100  that utilizes PFD  200   a  in locked status wherein charge pump current sources  114  and  116  are mismatched, in accordance with some embodiments of the present invention. For illustrative purposes, a situation wherein charge pump  112  sink current source  116  is larger than source current source  114  is considered. PLL  100  utilizing PFD  200   a , however, operates similarly when charge pump source current source  114  is larger than sink current source  116 . 
     PFD output signals UP  108  and DN  110  are asserted simultaneously for time period t 2 , corresponding to the cumulative delay time of AND gate  212 , delay buffer  216 , and the reset time of DFFs  206  and  208 . During this period, both sink current source  116  and source current source  114  inject current into PLL node  122 . As sink current source  116  is larger than source current source  114 , a negative net charge is injected into PLL node  122  when both current sources  114  and  116  are injecting charge. This net negative charge causes internal feedback clock signal  104  to lag reference clock signal  102  by fixed period t 1 . During period t 1 , source current source  114  injects charge into PLL node  122  to cancel the negative net charge injected into PLL node  122  during period t 2 . In this manner, a static phase error occurs between reference clock  102  and internal feedback clock  104  despite PLL  100  being in a phase locked status. 
       FIG. 4   a  illustrates a schematic diagram of a PFD that includes pulse blocking circuitry  400   a  in accordance with some embodiments of the present invention. PFD with pulse blocking circuitry  400   a  includes PFD  402  and pulse blocking circuitry  404 . PFD  402  includes two D flip-flops (“DFF”)  410  and  412 , AND gate  420 , and delay buffer  424 . Pulse blocking circuitry  404  includes two NAND gates  426  and  428  and two inverters  434  and  436 . 
     The D input of DFF  410  is coupled to input signal  406  which is set to a constant high logic level. Similarly, the D input of DFF  412  is coupled to input signal  408  which is set to a constant high logic level. In some embodiments, a single signal set to a constant high logic level may be coupled to the D inputs of both DFFs  410  and  412 . The clock input of DFF  410  is coupled to PLL reference clock signal  102 . Similarly, the clock input of DFF  412  is coupled to internal feedback clock signal  104  provided by VCO  126  of PLL system  100  via the feedback loop. DFF  410  Q output  416  and DFF  412  Q output  418  are provided as inputs to AND gate  420 . AND gate output  422  is delayed by delay buffer  424  and the delayed output is provided to DFFs  410  and  412  as reset signal  414 . 
     Q output  416  of DFF  410  is fed to one of the inputs of NAND gate  426 . The other input of NAND gate  426  is fed by the output  432  of NAND gate  428 . Similarly, Q output  418  of DFF  412  is fed to one of the inputs of NAND gate  428 . The other input of NAND gate  428  is fed by the output  430  of NAND gate  426 . NAND gate  426  output  430  is fed to inverter  434  which generates PFD  400   a  output signal UP  108 . NAND gate  428  output  432  is fed to inverter  436  which generates PFD  400   a  output signal DN  110 . 
       FIG. 4   b  illustrates an exemplary signal timing diagram  400   b  corresponding with PFD including pulse blocking circuit  400   a  in accordance with some embodiments of the present invention. The operation of PFD including pulse blocking circuitry  400   a  is described below with reference to signal timing diagram  400   b . For illustrative purposes, a situation wherein reference clock signal  102  leads internal feedback clock signal  104  is described. PFD  400   a , however, operates similarly when internal feedback clock signal  104  leads reference clock signal  102 . 
     The Q output  416  of DFF  410  captures the state of D input signal  406  at every rising edge of reference clock signal  102 . As DFF  410  D input signal  406  is set to a constant high logic level, at every rising edge of reference clock signal  102  DFF  410  Q output  416  is set to a high logic level after a period corresponding to the inherent capture delay time of DFF  410 . Similarly, the Q output  418  of DFF  412  captures the state of D input signal  408  at every rising edge of internal feedback clock signal  104 . As DFF  412  D input signal  408  is set to a constant high logic level, at every rising edge of internal feedback clock signal  104  DFF  412  Q output  418  is set to a high logic level after a period corresponding to the inherent capture delay time of DFF  412 . When Q output  416  of DFF  410  is set to a high logic level, output  430  of NAND gate  426  drops to a low logic level after a period corresponding to the delay time of NAND gate  426 . This in turn causes PFD  400   a  output UP  108  to be set to a high logic level. Further, this keeps output  432  of NAND gate  428  set to a high logic level thereby preventing output DN  110  from being asserted. 
     Once Q output  416  of DFF  410  and Q output  418  of DFF  412  are both set to a high logic level, resetting of DFF  410  and DFF  418  initiates. AND gate  420  output  422  is set to a high logic level after a period corresponding to the inherent delay time of AND gate  420 . AND gate  420  output  422  is delayed by delay buffer  424  and the delayed AND gate output is provided to DFF  410  and DFF  412  as reset signal  414 . Once reset signal  414  is asserted and after the inherent reset delay time of the DFFs  408  and  410 , DFF  410  Q output  416  and DFF  412  Q output  418  reset to a low logic level. DFF  410  Q output  416  and DFF  412  Q output  418  remain at this level until DFF  410  and DFF  416  Q outputs  416  and  418  capture inputs  406  and  408  at the next rising clock edges of reference clock signal  102  and internal feedback clock signal  104  respectively. After Q outputs  416  and  418  reset to a low logic level, NAND gate  426  output  430  is reset to a high logic level, thereby causing PLL output UP  108  to also reset to a low logic level. 
     When reference clock signal  102  leads feedback clock signal  104 , only PFD output signal UP  108  is asserted. Similarly, when reference clock signal  102  lags feedback clock signal  104 , only PFD  400   a  output signal DN  110  is asserted. By preventing PFD output signals UP  108  and DN  110  from simultaneously being set to a high logic level, the PFD utilizing pulse blocking circuitry  400   a  shown in  FIG. 4   a  eliminates the effects of charge pump  112  current source mismatch and significantly reduces static phase errors. 
       FIG. 5  illustrates an exemplary signal timing diagram  500  of PLL  100  in locked status utilizing a PFD that includes pulse blocking circuitry  400   a , in accordance with some embodiments of the present invention. As shown in  FIG. 5 , PFD output signals UP  108  and DN  110  are not asserted simultaneously by PFD with pulse blocking circuitry  400   a . Accordingly, only one of current sources  114  and  116  of PLL charge pump  112  injects charge at a given time, thereby significantly reducing any static phase error caused by mismatch of charge pump  112  current sources  114  and  116 . The net charge injected  502  by PLL charge pump  112  has equal magnitude but opposite polarities in every successive two periods, resulting in zero average net injected charge. Reference clock signal  102  and PLL internal feedback clock signal  104  lead each other alternatively by a period corresponding to remaining non-ideal phase shift t 3 . Non-ideal phase shift t 3 , attributed to static phase error between reference clock signal  102  and PLL internal feedback clock signal  104 , is greatly minimized. In some embodiments, non-ideal phase shift t 3  may be as small as sub-picoseconds. 
     Other embodiments of the invention will be apparent to those skilled in the art from consideration of the specification and practice of the invention disclosed herein. It is intended that the specification and examples be considered as exemplary only, therefore, the invention is limited only by the following claims.