Abstract:
A feedback enhanced triggering device for an electrostatic discharge protection circuit includes: a first inverter  30   b  having an output coupled to an input of a second inverter  30   c , the second inverter  30   c  having an output coupled to a control node for a discharge device  31  such as a transistor; a high side feedback transistor  34  coupled to the output of the first inverter  30   b , and having a control node coupled to the output of the second inverter  30   c ; and a low side feedback transistor  35  coupled to the output of the first inverter  30   b , and having a control node coupled to the output of the second inverter  30   c , wherein the feedback transistors  34  and  35  provide enhanced triggering for electrostatic discharge protection.

Description:
FIELD OF THE INVENTION  
         [0001]    This invention generally relates to electronic systems and in particular it relates to electrostatic discharge protection circuits.  
         BACKGROUND OF THE INVENTION  
         [0002]    [0002]FIG. 1 depicts a circuit diagram of a portion of an electrostatic discharge (ESD) protection circuit  10  that contains an Input/Output (I/O) pad  20 . The I/O pad  20  further contains bondpad  24 , diodes  25  and  26 , and MOSFETS  27  and  28 . Within this specification the first current electrode of MOSFET devices is called the source terminal, which is coupled to one of the power supply terminals VDD or VSS unless otherwise specified. The second current electrode of those same MOSFET devices is called the drain terminal. The drain terminal of MOSFET  28  is coupled to bondpad  24 , as is the drain terminal of MOSFET  27 . In I/O pad  20 , the anode of diode  26  is coupled to bondpad  24  and the cathode of diode  26  is coupled to supply terminal VDD. Similarly, the cathode of diode  25  is coupled to the bondpad  24  and the anode of diode  25  coupled to terminal VSS. The ESD protection circuit  10  further contains a rail clamp  30  with a first current electrode coupled to supply terminal VDD and a second current electrode coupled supply terminal VSS. ESD protection circuit  10  also contains a diode  40  where the anode of diode  40  is coupled to supply rail VSS and the cathode of diode  40  is coupled to supply terminal VDD. The ESD protection circuit  10  also contains parasitic bus resistances PP and RG in the supply lines. These are not explicit resistor elements, but rather ones that exist by default due to the inherent resistance of any electrical conductor. In general, the greater the distance between the I/O pad  20  and the ESD rail clamp  30  and bus diode  40 , the greater the value of the parasitic resistances RP and RG.  
           [0003]    Integrated circuits must be protected against electrostatic discharges in order to prevent permanent damage that can impair or eliminate desired functionality. ESD damage normally occurs in the MOSFET devices or interconnecting layers used to couple MOSFETs together to form a circuit. Each pin in an integrated circuit must be coupled to an appropriate ESD protection circuit such that the ESD discharge current is shunted away from the internal portions of the chip that are the most sensitive to damage. As such, ESD discharge paths must be provided between every pair of pins in an IC for both positive and negative polarities.  
           [0004]    The function of diode  26  in FIG. 1 is to provide a shunting path to the rail VDD for ESD currents produced by ESD potentials applied to bondpad  24  which are significantly more positive than anywhere else on the IC. Similarly, the function of diode  25  is to provide a shunting path for ESD currents that are produced by ESD potentials that are significantly more negative than elsewhere on the IC. The function of rail clamp  30  is to provide a coupling between the rails VDD and VSS for those ESD paths that require such a coupling in order to complete the discharge loop. For example, for pad positive-to-VSS stress, where bondpad  24  is taken positive with respect to the rail VSS, the ESD discharge current  100  flows from bondpad  24 , through diode  26 , along the rail VDD (through the parasitic resistor RP) and back to the rail VSS through rail clamp  30 . In general, the goal is to keep the maximum voltage built-up in the discharge loop to within acceptable limits. Similarly, for pad positive-to-VDD, where the bondpad  24  is taken positive relative to the rail VDD, discharge current  102  flows from bondpad  24 , through diode  26  and to the rail VDD where the circuit is completed.  
           [0005]    For negative ESD events applied to bondpad  24  such as pad negative-to-VSS stress, ESD current  101  flows from the rail VSS (which is now positive relative to the bondpad  24 ) through diode  25  and back to bondpad  24  where the circuit is completed. Finally, for pad negative-to-VDD, the ESD current  103  flows from the rail VDD, through rail clamp  30 , along the rail VSS (through parasitic resistance RG) and back to bondpad  24  through diode  25 . Again, the goal is to keep the maximum voltage built-up in the discharge loop to within acceptable limits. In each case described so far, diode  25  or  26  alone or a combination of diode  25  or  26  and rail clamp  30 , act to provide a shunting path for the ESD current such that these currents are kept from the sensitive internal portions of the chip. For ESD stress applied to bondpad  24 , output buffer transistors  27  and  28  are the most susceptible to damage and so is any other input buffer circuitry (not shown) coupled to bondpad  24 .  
           [0006]    Just as ESD pulses can be applied between the I/O pads and the supply rails, ESD discharges can occur between the power supply rails. For example for rail VDD positive-to-VSS stress, ESD current  104  flows through the rail clamp  30  from the rail VDD to the rail VSS. For rail VSS positive-to-VDD stress, ESD current  105  flows from the rail VSS, through diode  40  and to the rail VDD. Thus, the rail clamp circuit is a fundamental component in providing a discharge path for ESD polarities (positive or negative) which cause the first current electrode of the rail clamp to be more positive than its second current electrode. Although the several ESD discharge paths shown in FIG. 1, there are other paths not shown between other pairs of pins which are well known to one skilled in the art. These have not been described here simply for brevity and do not detract in any way from the description or understanding of the invention described herein.  
           [0007]    ESD discharges are brief transient events that are usually less than one microsecond in duration. Furthermore, the rise times associated with these brief pulses are usually less than roughly twenty nanoseconds. When ESD pulses are applied to the I/O pads of a chip, they produce similar brief, fast rise time potentials on the power supply rails due to the presence of ESD protection diodes  25  and  26  in FIG. 1. Thus, the rail clamp circuit must be able to detect these fast transients and begin conducting so as to shunt the resulting ESD current. However, the rail clamp must not respond to the much slower rise times (greater than 1 millisecond) which are present on the power supply rails during normal power-up events in regular chip operation. If the ESD rail clamp were to trigger and conduct during normal power-up events, the desired operation of the IC could be compromised. Furthermore, in addition to triggering when needed for ESD protection, the rail clamp circuits must stay in a highly conductive state for the entire duration of the ESD pulse so that all of the ESD energy is safely discharged. If the rail clamp circuit were to shut-off prematurely, damaging potentials would build-up quickly between the power rails and cause device failure.  
           [0008]    [0008]FIG. 2 depicts a wider portion of an integrated circuit that shows how rail clamp  30  can be placed relative to the I/O pads it is protecting. Here, rail clamp  30  has been placed in the VSS pads in the chip that are responsible for supplying power connections for the IC. The rail clamp  30  can also be placed in VDD pads. These placements are shown as  30 L 1  and  30 R 1 . In this manner, many I/O cells ( 20 L 2 ,  20 L 1 ,  20 ,  20 R 1 ,  20 R 2 ) are able to share ESD rail clamp  30 L 1  and  30 R 1  that results in more robust ESD protection and reduced chip area. Alternatively, the size of an individual rail clamp can be reduced if more than one can be relied upon to conduct ESD current that also saves die area. In general, the sum total of parasitic power and ground rail resistances (RPL 2 , RPL 1  RP RPR 1 , RPR 2 ) and (RGL 2 , RGL 1  RG RGR 1 , RGR 2 ) around the ESD discharge loop sets the limit on how far apart ESD rail clamps can be spaced in order to achieve a given level of ESD protection. The overall goal is to keep the maximum voltage that occurs at the bondpad during an ESD discharge within acceptable limits so that damage does not occur in the sensitive circuit elements.  
           [0009]    In an effort to mitigate the effects of parasitic bus resistance, one may distribute the ESD rail clamps locally in the I/O cells themselves ( 20 BL 1 ,  20 B,  20 BR 1 ). This is shown in FIG. 3 where several smaller local clamps are used where one is placed in each I/O pad. In this manner, several ESD rail clamps must now participate in the ESD event to achieve robust protection but the effects of power and ground rail resistances may be reduced over locating the larger clamps in more centralized locations. In general one skilled in the art is able to make a determination as to whether the scheme described in FIG. 3 or in FIG. 2 is more applicable to their particular application. This in no way limits the general usage of the ESD rail clamp described herein, as one skilled in the art is able to easily scale the relative sizes of the clamp so that an optimal tradeoff is achieved.  
         SUMMARY OF THE INVENTION  
         [0010]    A feedback enhanced triggering device for an electrostatic discharge protection circuit includes: a first inverter having an output coupled to an input of a second inverter; the second inverter having an output coupled to a control node for a discharge device such as a transistor; a high side feedback transistor coupled to the output of the first inverter, and having a control node coupled to the output of the second inverter; and a low side feedback transistor coupled to the output of the first inverter, and having a control node coupled to the output of the second inverter, wherein the feedback transistors provide enhanced triggering for electrostatic discharge protection. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0011]    In the drawings:  
         [0012]    [0012]FIG. 1 is a schematic circuit diagram of a portion of an electrostatic discharge (ESD) protection circuit;  
         [0013]    [0013]FIG. 2 is a schematic circuit diagram of a portion of an integrated circuit that shows the placement of ESD protection circuits relative to the I/O pads that are protected;  
         [0014]    [0014]FIG. 3 is a schematic circuit diagram of a portion of an integrated circuit that shows the ESD protection circuits placed locally in the I/O cells;  
         [0015]    [0015]FIG. 4 is a schematic circuit diagram of an ESD protection circuit with feedback enhanced triggering according to the present invention;  
         [0016]    [0016]FIG. 5 is a plot of the internal node voltages in the circuit of FIG. 4 during a 4 kV Human Body Model (HBM) ESD discharge;  
         [0017]    [0017]FIG. 6 is a plot of individual transistor currents in the circuit of FIG. 4 during an ESD discharge;  
         [0018]    [0018]FIG. 7 is a plot of the current in the ESD discharge transistor of FIG. 4 during an ESD discharge;  
         [0019]    [0019]FIG. 8 is a plot of the internal node voltages in the circuit of FIG. 4 during power up and normal operation;  
         [0020]    [0020]FIG. 9 is a plot of individual transistor currents in the circuit of FIG. 4 during power up and normal operation;  
         [0021]    [0021]FIG. 10 is a plot of the current in the ESD discharge transistor of FIG. 4 during power up and normal operation.  
     
    
     DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS  
       [0022]    [0022]FIG. 4 depicts a schematic diagram of the rail clamp circuit of this invention. The circuit is comprised of an RC circuit  30   a  (timing circuit), a first inverter circuit  30   b  (CMOS inverter), a second inverter circuit  30   c  (CMOS inverter), an ESD transistor  31  (ESD device), a feedback NMOS device  35  and a feedback PMOS device  34 . The RC circuit  30   a , is further comprised of PMOS transistor  38  that functions as a capacitor and NMOS transistor  39  which functions as a resistor. PMOS transistor  38  has its first and second current electrodes, and its well electrode coupled to the VDD supply rail. The control electrode of transistor  38  is coupled to node RC. The NMOS transistor  39  has its source electrode coupled to the VSS rail and its drain electrode coupled to node RC. The control electrode of transistor  39  is coupled to the node VDD. Inverter  30   b  is comprised of PMOS transistor  36  and NMOS transistor  37 . PMOS transistor  36  has its source and well terminal coupled to the rail VDD, its drain terminal coupled to node INV 1 -OUT and its control electrode coupled to node RC. NMOS transistor  37  has its source electrode coupled to node VSS, its drain electrode coupled to node INV-OUT 1  and its control electrode coupled to node RC. Similarly, Inverter  30   c  is comprised of PMOS transistor  32  and NMOS transistor  33 . PMOS transistor  32  has its source and well terminal coupled to the rail VDD, its drain terminal coupled to node BIG-GATE and its control electrode coupled to node INV 1 -OUT. NMOS transistor  33  has its source electrode coupled to node VSS, its drain electrode coupled to node BIG-GATE and its control electrode coupled to node INV 1 -OUT. ESD discharge transistor  31  has its drain terminal coupled to the rail VDD, its source terminal coupled to the rail VSS, and its control electrode coupled to node BIG-GATE. PMOS feedback transistor  34  has its source and well terminals coupled to the rail VDD and its drain terminal coupled to node INV 1 -OUT. The control electrode of transistor  34  is coupled to node BIG-GATE. Finally, NMOS feedback transistor  35  has its source electrode coupled to the rail VSS, its drain electrode coupled to node INV 1 _OUT and its control electrode coupled to node BIG-GATE.  
         [0023]    The operation of the circuit shown in FIG. 4 under ESD conditions will now be described. Prior to the ESD event, the integrated circuit is not energized and all node voltages can be considered at zero potential. A fast positive going ESD transient on the power rail, causes node RC to rise instantaneously along with the VDD potential due to the displacement current flow in the MOSFET capacitor  38 . The elevation of node RC causes transistor  37  to be placed into a conductive state which in turn pulls the node INV 1 -OUT towards the ground potential VSS. This in turn causes transistor  32  to be placed into a conductive state which then couples the control electrode of ESD discharge transistor  31  towards the VDD potential. Thus, transistor  31  is now placed in a conductive state and is now free to begin to shunt the ESD current. Once the potential of node BIG-GATE has risen to a threshold potential above the rail VSS, NMOS feedback transistor  35  begins to conduct. Current conduction in transistor  35  further pulls the potential of node INV 1 -OUT towards ground, which further enhances current conduction in transistor  32 , which then pulls the potential of BIG-GATE closer to that of the rail VDD. In the limit, the potential of node INV 1 _OUT is at ground VSS and the potential of node BIG-GATE is identical to that of the rail VDD which ensures that ESD discharge transistor  31  is conducting as strongly as possible. This completes a feedback loop, which “latches” transistor  31  into a conductive state.  
         [0024]    Once transistor  31  has been latched into a conductive state, the time constant of the RC circuit  30   a  is now free to time out. This is highly desirable since this means that the duration of this time constant can be significantly shorter than the ESD event which translates into an RC network with greatly reduced physical area. While the rail clamp circuit is transitioning into this conductive state, PMOS feedback transistor  34  will actually impede the collapse on node INV 1 -OUT to ground, as long as the potential of node BIG-GATE is near ground prior to the full turn-on of transistor  31 . This is actually a highly desirable effect since it functions as a mechanism by which to prevent false triggering. As the potential on node BIG-GATE begins to rise, transistor  34  begins to shut-off which then accelerates the collapse of node INV 1 -OUT which in turn elevates node BIG-GATE via enhanced conduction in transistor  32 . One skilled in the art can adjust the dynamic current balance of transistor  37 , transistor  34 , and feedback transistor  35  to achieve a minimum critical voltage needed on the rail VDD to trigger the rail clamp once the transient change on the rail VDD has caused the initial action. This is a highly novel and desirable effect since false triggering can be controlled.  
         [0025]    The timing out of RC circuit  30   a  means that NMOS resistor  39  has had enough time to discharge the potential on node RC towards ground VSS. This in turn causes PMOS device  36  to begin to conduct. This action now tries to elevate the potential of node INV 1 -OUT towards the rail VDD which conversely attempts to turn-off the clamp. The potential on node INV 1 -OUT is set by the current balance between transistor  36  and transistor  35 . The settling potential of this node is set such that the clamp stays on until a minimum critical residual energy is reached in the ESD pulse. Once the energy in the ESD pulse has reached this critical point, the latching action of the circuit will release and transistor  31  will again be placed into a non-conductive state. By this time, the ESD pulse has either dissipated or does not have sufficient energy to cause damage to the integrated circuit.  
         [0026]    [0026]FIG. 5 depicts a SPICE simulation of the internal node voltages in rail clamp  30  during a 4 kV Human Body Model (HBM) ESD discharge. Here a positive-to-VSS HBM pulse is applied between the VDD and VSS terminals of the circuit. In FIG. 5, the potential of node RC is shown to instantaneously respond to the quick rise of the VDD rail and later, after some time, release back to its resting value. This is the triggering and time-out of the RC circuit  30   a . Similarly, node INV 1 -OUT is shown to initially drop to and remain at a potential near ground. As stated previously, initial coupling to ground of node INV 1 -OUT is caused by turn-on of transistor  37  and the subsequent holding of node INV 1 -OUT at ground (after RC timeout) results from feedback NMOS transistor  35 . Node BIG-GATE is shown to be coupled to the rail VDD which results in ESD discharge transistor  31  being placed in a highly conductive state.  
         [0027]    [0027]FIG. 6 shows the individual transistor currents in device  36  and  37  of Inverter  1  ( 30   b ) and feedback NMOS transistor  35  and feedback PMOS transistor  34 . The aggregate effect of these currents determines the voltage profile exhibited by node INV 1 -OUT. FIG. 6 shows transistor  37  as initially conducting current due to turn-on caused by node RC and current flow in transistor  36  being essentially zero. Once the RC time constant begins to turn-off, current flow in transistor  37  diminishes while current flow in transistor  36  increases and remains present for the majority of the ESD event. Similarly, current flow in NMOS feedback device  35  begins instantly as does that in PMOS feedback device  34 . However, the current flow in transistor  34  soon disappears due to the charging of node BIG-GATE. Thus the maintenance of the voltage of node INV 1 -OUT over the remainder of the ESD pulse is due to the current balance between transistor  36  and  35 .  
         [0028]    [0028]FIG. 7 depicts the current flow between the two current electrodes of ESD discharge transistor  31 . It can be seen here that transistor  31  conducts a large (2.6 Amp) ESD current. This necessitates that transistor be a relatively large device to conduct this magnitude of current. Typically, transistor  31  is in the range of 1000 um to 3000 um in total width depending on the number of rail clamps, which are expected to participate in the ESD event in accordance with FIG. 2. If smaller local clamps are used as in the case in FIG. 3, then less total width can be used for each individual clamp  30 . One skilled in the art will be able to determine the optimal sizing of rail clamp  30  for a given application and usage.  
         [0029]    In general it is important that the ESD rail clamp remain in a non-conductive state during system power-up and normal operation. In normal applications during power-up, the system power supply will ramp at a predefined rate, which is usually in the range of several milliseconds to several tens of milliseconds. This is orders of magnitude slower than the rise times seen during ESD events. As before, we will assume that the chip is unpowered and that all internal node voltages are at essentially zero volts. In response to the slowly increasing voltage rate on the rail VDD, the RC node in FIG. 4 remains at a potential near ground since the NMOS resistor  39  can effectively remove any displacement charge deposited by PMOS capacitor  38 . This displacement current is minimal since the dV/dT of the power rail is low. If node RC remains near ground then node INV 1 -OUT remains at the instantaneous potential of the rail VDD via current conduction in transistor  36 . Since the node INV 1 -OUT is essentially at the potential of the rail VDD, node BIG-GATE is coupled to ground VSS due to current conduction in transistor  33 . This maintains ESD discharge transistor  31  in a non-conductive state. Feedback transistor  34  will also be in a conductive state which further couples node INV 1 -OUT to the rail VDD which is highly beneficial. It will be shown next that feedback transistor  34  has an important role to play in preventing false triggering.  
         [0030]    While most system power supplies ramp in a highly controlled and deterministic manner, there are certain applications where faster than normal power supply transitions occur. One such example is where a system board must be “hot plugged” where it is not practical to power down the entire system. In this case, the transient that results to the individual chip power supplies on the board can be significantly faster than normal. The ESD protection network must not respond to this transient. This can be seen in FIG. 8 where a very fast power supply ramp time of 1 microsecond is applied. Here node RC is elevated to a potential of roughly 0.75V, which is enough to initiate current conduction in transistor  37 . This momentarily stalls the rise of node INV 1 -OUT. Recall that if node INV 1 -OUT is allowed to reach a potential closer to ground, triggering of the rail clamp  30  will result. FIG. 9 depicts the currents in devices  34 ,  35 ,  36  and  37 , which determine the voltage of node INV 1 -OUT. Here we see the current conduction in transistor  37  but we also see that PMOS feedback transistor  34  is opposing this current flow. The net result is that node INV 1 -OUT soon returns to its tracking of the ramping of the rail VDD and the clamp is prevented from false triggering. FIG. 10 depicts the current flow in the ESD discharge transistor  31  which shows that a minimal value of current flows in this device during the intermediate state and that this value then decays to zero after the transient has resolved. Also notice (FIG. 8) that once the RC time constant has timed out, current flow in transistor  37  soon decays. If a much longer time constant were needed for ESD operation, then there would be a greater risk of triggering the clamp for a fast ramp time of the rail VDD, as transistor  37  would be trying to keep node INV 1 -OUT at ground for a longer period of time. Thus, the combination of a short time constant and feedback transistor  34  allows for very fast power supply ramp times to be used which is a very highly desirable feature of the invention described herein.  
         [0031]    While this invention has been described with reference to an illustrative embodiment, this description is not intended to be construed in a limiting sense. Various modifications and combinations of the illustrative embodiment, as well as other embodiments of the invention, will be apparent to persons skilled in the art upon reference to the description. It is therefore intended that the appended claims encompass any such modifications or embodiments.