Abstract:
The invention in general relates to Medical Instruments for bone fracture healing, ultrasonic surgery, tissue ablation and cutting and drilling, dry powder inhalers and more particularly to a method to monitor load condition and overall efficiency, safety and reliability of ultrasonic energy delivery using a Class E power Amplifier. Three parameters: frequency (f), duty cycle (D) and Power Amplifier supply voltage (Vdd) are continuously adjusted in order to maintain optimal and suboptimal Class E operation. The load for the power driver can be a single element or a stack. For every frequency and duty cycle of operation the MOSFET drain voltage is compared to a known value which is proportional to Power Amplifier supply voltage Vdd. During the MOSFET OFF part of operation a timer is started when drain voltage falls below a known threshold value. The timer is stopped by the MOSFET leading edge gate clock. A measure of the distance from optimal operation is implemented in the form of a counter value N thus allowing a continuous load monitoring. A one Class E Amplifier clock cycle reaction to fast load change is thus possible and therefore a robust efficient operation, with increased safety and reliability of Ultrasonic power delivery.

Description:
BACKGROUND OF THE INVENTION 
     In Medical Instrumentation applications using ultrasonic energy as means to perform work, many solutions have been presented for different constitutive building blocks like current/voltage feedback loops, driver Amplifier (single stage, push-pull, half bridge, full bridge). In bone fracture healing and various surgical Instruments a wide load variation is expected and for safety and reliability reasons a fast response to fast load change is desired. These Instruments use piezoelectric single elements or stacks as load for the power amplifier. For low acoustic power application, in general less than 5 Watts a single element is used, for higher power, for example higher than 20 Watts a stack of elements is preferred. The electrical parameters [8], [9] are well defined by the manufacturer and they depend on the mechanical action that they have to perform. The mechanical work varies widely and due to the interdependence between electrical and mechanical parameters, the Power Amplifier driving the piezoelectric load “sees” a continuously varying electrical load. Fast reaction to fast load change is therefore something desirable. Typical piezoelectric constructions in Medical applications can be exemplified in U.S. Pat. No. 4,530,138 or U.S. Pat. No. 3,889,166 
     1. Field of the Invention 
     The invention in general relates to Medical Instruments delivering ultrasonic energy to perform work and more particularly to improvements related to load monitoring, by implementing a method to digitally measure the distance from optimal load conditions and also a method to fast react (and shut down if necessary) due to load changes in one clock cycle of the Power Amplifier if the load changes in one cycle. 
     2. Description of Prior Art 
     For cutting and cauterization applications, In Surgical Instrument presented in U.S. Pat. No. 7,273,483 a solution is offered for an environment where transducer temperature increase can lead to material fatigue and possible failure. Mainly for adjusting Power to the load via a push-pull amplifier, it measures load conditions via current averaging and voltage averaging followed by analog to digital (ADC) conversion in a feedback loop control scheme. Reaction time to load change is thus determined by current and voltage averaging filters plus ADC conversion time, too slow for a sudden load change which could lead to Instrument failure. A similar feedback approach is adopted in U.S. Pat. Nos. 5,026,387 and 4,056,761. These Instruments cannot react fast enough to prevent a catastrophic load condition which could lead to shatter or breakage. 
     In Exogen 2000+SAFHS, a Pulsed Ultrasound Instrument for bone fracture healing manufactured by Smith and Nephew located in Memphis, Tenn., a generic averaging filter is used to determine transducer electrical impedance variation due to the amount of coupling gel used and thus determining gel/no gel alarm conditions (U.S. Pat. No. 6,261,249). These Pulsed Ultrasound Instruments under sudden load change can also lead to Instrument failure. Therefore there is a need for a fast Instrument response due to fast load change. In this application a Class E Power Amplifier driving a piezoelectric load is used as Power Stage. Class E amplifier is well known. Fast feedback Class E operation is also not new. A generic Class E amplifier as final power stage and a predictor to load change was presented in [1]. It was tested by slightly adjusting the frequency of operation. In reference [1] a resistor is placed in series with the source of power transistor Q 1  and an additional circuit is built to indicate class E or non class E operation, however such a circuit has a time response of 20 to 30 ms when operating at 1 MHz which in certain situations like highly load sensitive ultrasonic surgery instruments and Pulsed Ultrasound can be too slow. FIG. 4A in U.S. Pat. No. 3,919,656 and reference [1]  FIG. 2  present typical Class E waveforms under variable load quality factor (Q). Design of Class E power Amplifiers under nominal or off nominal (variable duty cycle D, or duty ratio as it is considered in [2], [3]) is also known. 
     SUMMARY OF THE INVENTION 
     The present invention presents a method to control the Ultrasonic power delivered to a piezoelectric or magnetostrictive load using a Class E Amplifier by means of digitally monitoring in each cycle the amplifier operation. This Power Amplifier is first designed so that it presents a predefined sensitivity to load changes and these changes are monitored in each clock cycle by means of a timer content. A trend can therefore be determined as successive values of a counter timer number. An emergency shut down is thus possible in one clock cycle of the first occurrence of non optimal operation of the Class E amplifier due to load change. 
     A non optimal operation is a non-zero MOSFET drain voltage at MOSFET turn on. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a general illustration of an ultrasonic energy delivery method and apparatus for high efficiency, high safety in accordance with an exemplary embodiment 
         FIG. 2  is an illustration of optimal Class E drain voltage landing and the start-stop timer mechanism. 
         FIG. 3  is an illustration of a suboptimal Class E drain voltage landing, due to a higher Q where the timer content is higher due to the fact that time difference t 2 -t 1  is higher than in optimal case. 
         FIG. 4  represents a lower than nominal Q load variation and the MOSFET drain voltage landing. Moment t 1  is very close to t 2  and the timer content is zero. 
         FIG. 5  is an illustration of the software loop decision blocks to apply or not MOSFET gate pulse in the next cycle (period). 
         FIG. 6  shows a dynamic transition from optimal Class E operation to non Class E operation due to fast load change and the fast action of the feedback loop in the present method. 
         FIG. 7  graphically illustrates the efficiency indicator (i.e. timer content) for different modes of Class E operation. 
     
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENT 
     In reference to  FIGS. 1 to 7  a description of the preferred embodiment is presented. 
     Ref to  FIG. 1 , a microcontroller MCU  100  is used to program the programmable oscillator  106  via an interface  105 . Examples of such components could be Texas Instruments, Dallas Tex., MSP430FG437 for the MCU and Linear Technology, Milpitas Ca, LTC6904 for the frequency f programmable oscillator. The digital signal from 106 is gated with the processor logic signal  112  in gate  113  in order to generate bursts of pulses according to treatment indication or a continuous digital signal. For example the duration of the logic “1” signal  112  could be 200 μs followed by 800 μs “0” logic for a 1 KHz PRF (Pulse Repetition Rate) ultrasound delivery system, or a continuous digital logic “1” for 25 KHz to 100 KHz Ultrasonic surgery Instruments. R 9 , C 5  is an integrator circuit, the resultant triangular pulses are applied to the inverting input of comparator  115 . The non-inverting input of this comparator takes a continuously variable voltage signal from a digital to analog converter, D/A on line  121  via resistors R 7 , R 8 , R 11  and as a result the digital signal at the output of the gate  114  is adjustable both in duty cycle D, and frequency f. This is block  117  shown in an accolade, the waveform is shown pictorially in block  118 . The driving signal is applied to MOSFET Q 1  gate via the coupling capacitor C 4  and resistor R 5  in junction  126 . This digital signal also drives, via connection  127  the clock input of the D flip-flop  102 . Another comparator,  116  takes at the inverting input a fraction (k) of the MOSFET drain voltage via resistor divider R 1  and R 2  and at the non-inverting input the power Amplifier voltage Vdd, via resistors R 3  and R 4 . Resistors R 1  and R 3  are connected to ground at line  132 . This voltage, Vdd is generated by the DC/DC converter  101 , preferably a high speed Class E converter. Such a converter is known for one skilled in the art and has the advantage of having both step-up and step-down capabilities [4], [6]. For fast DC/DC ON/OFF time, the switching frequency of the converter  101  should be at least 10 times higher than the frequency generated by the oscillator  106 . For example, for f=1.5 MHz used in bone fracture healing Instruments the frequency generated by the processor  100  on line  108  should be at least 15 MHz to start the DC/DC converter  101 . For 20 KHz to 100 kHz Ultrasonic surgery Instruments, at least a 1 MHz frequency Class E DC/DC is required. The Enable line  109  is a logic signal to turn ON/OFF the converter. R 10  and C 7  provide a low pass filter function for comparator  116  output and is connected, via line  128  to the D input of the D flip-flop  102 . The input D of the flip-flop  102  is also connected via line  134  to an MCU  100  timer input. The inverted output  107  (Ō) is connected to an MCU (or processor)  100  input interrupt line INT 1 . The fast dynamic action on INT 1  input can be explained with reference to  FIG. 6  where, due to a fast load change, a transition from optimal Class E to non-optimal (non Class E) drain voltage operation is illustrated. As soon as, due to drain voltage change, the comparator output waveform  605  trailing edge falls behind the trailing edge of the Mosfet Q 1  gate voltage  603 , the interrupt line  607  normally at “0” logic is now fast switched to logic 1 thus providing an interrupt INT 1  to the processor  100 . It is recognized here that this constitutes a fast feedback mechanism for class E operation with one Mosfet clock cycle delay. The processor can now decide whether to stop providing further pulses and shut down operation due to abnormal load conditions or continue providing pulses and monitor more closely the load via Mosfet drain voltage. 
     Vcomp and V threshold in  FIG. 6  represent the input voltages of comparator  116 . Ref. now to  FIG. 2 , an optimal Class E drain voltage is shown. The horizontal dotted line represents ground reference voltage (GND) as in  FIGS. 3 and 4 . D represents the Mosfet turned OFF portion of a typical clock cycle. In normal load conditions (optimal or nominal operation as it is known in published literature, for example ref. [2], [3] the drain voltage during the Mosfet turn-on (t 2 ) should be zero (“0”) with “0” slope. 
     The comparator  116  V threshold  voltage is such that the drain voltage falls below this level at moment t 1  and at this moment the comparator output switches to logic 1, starting the timer counter. When the Mosfet turns on at moment t 2  the software stops the counter and the count number N indicates a typical class E optimal drain voltage landing. The falling edge of the Mosfet clock at moment t 3  happens before the comparator output falling edge at moment t 4  due to low pass filtering action of R 10  and C 7 . So in normal (optimal) operation, in each MOSFET clock cycle a timer is started by comparator  116  output connection  134  and is stopped by software via connection  135 , which is another interrupt line (INT 2 ) to the processor. INT 1  has higher priority than INT 2  since it indicates a non Class E operation. The timer content, N gives thus an indication of Class E Power Amplifier distance from optimal operation. In this load condition the D flip-flop output stays always “0” logic since the clock for D flip-flop  102  always “sees” a “1” (and latches it) at the D input and therefore the negated output is “0”. The counter start signal always occurs before the Mosfet clock arrives, therefore no Interrupt INT 1  is triggered. At the moment t 2  the count is N 1 ±ΔN as illustrated in  FIG. 7 . 
     A sub optimal class E drain voltage landing is shown in  FIG. 3  where the drain voltage falls to zero with non-zero slope. The drain voltage is limited to almost zero (actually slightly negative due to the Mosfet body diode). The distance between moments t 1  and t 2  is higher and the content N 2  of the counter is higher than previously and thus the timer content (see N 2  in  FIG. 7  also) provides a digital indication of the distance from optimal Class E operating condition. Moments t 3  and t 4  occur as in  FIG. 2  and D flip-flop output remains “0” and no interrupt is triggered. Optimal ( FIG. 1 ) and suboptimal (fig 2 ) conditions in Class E Amplifier operation are considered high efficiency because the voltage across MOSFET is zero at turn on (moment t 2 ). 
     In  FIG. 4  a non-ZVS (non Zero Voltage Switching) drain voltage landing is shown, where moment t 2  occurs before t 1 . In this case comparator output remains “0” when the Mosfet clock trailing edge is applied and thus it latches logic “1” on Interrupt INT 1 , line  107 . So comparator  102  output is now “0” and D flip-flop output is “1”. Fast load conditions will lead to fast drain voltage change and as explained above, this is illustrated in  FIG. 6  Here it is seen that as soon as the leading edge of comparator  116  signal output  605  falls behind Mosfet clock cycle at moment  608 , an interrupt is triggered by the leading edge of the MOSFET gate pulse  608  due to a gradual transition from optimal to non optimal conditions and this change on D flip-flop output  606  is as fast as the load change. This is where “one clock cycle” shutdown is obvious. As soon as this condition occurs, the processor can stop providing pulses in the next cycle, so the abnormal load condition time resolution is one clock cycle and the condition can be addresses in the next clock cycle. 
       FIG. 5  represents a software flowchart diagram explaining the decision to further apply or not pulses based upon the Timer value N. N=N 1 ±ΔN indicates an optimal operating condition ( FIG. 7 ). If the value has a tendency to increase from cycle to cycle this is an indication of transition from optimal to suboptimal Class E operation (N 2 ) and pulses are applied normally. If the trend from cycle to cycle is towards N decreasing, this is an indication that a non Class E operation is imminent and if the timer content falls below a predetermined safety value then the processor stops applying pulses by bringing line  112  to gate  114  to “0” logic. 
     In what follows, the analytical value of the Mosfet Q 1  drain voltage is calculated along with V threshold  and V comparator . 
     This value depends on the duty cycle D, frequency f, power stage supply voltage Vdd and load phase angle φ. 
     The analysis is carried out with the usual assumptions for Class E operation [1]-[5] and considering load  131  in  FIG. 1  connected at point  110  as pure resistive, with value R. 
     From  FIG. 1  with angular switching frequency w=2πf and θ=ωt and considering the R 1 +R 2  resistor value large enough so the current through it can be ignored, the inductor L 1  current Idd is:
 
 Idd=I   c   +I   0   (1)
 
where I c  is the current through the capacitor C 1  and I 0  is the load current.
 
 I   c   =i   c (θ), I   0   =I   m  sin(θ+φ)  (2)
 
 i   c (θ)= Idd−I   m  sin(θ+φ)  (3)
 
     The Mosfet drain voltage is the voltage across Capacitor C 1 , which is 
     
       
         
           
             
               
                 
                   
                     
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     With zero voltage condition at Mosfet turn on
 
 Vs (2 πD )=0  (6)
 
equation (5) becomes
 
     
       
         
           
             
               
                 
                   
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     Considering ideal components, the load output power is equal to the input power, i.e.
 
 IddVdd=I   m   2   R/ 2  (8)
 
so substituting Idd from (8) in (7), the load current amplitude is
 
                     I   m     =       2   ⁢           ⁢   V   ⁢           ⁢   d   ⁢           ⁢   d   ⁢           ⁢   sin   ⁢           ⁢   π   ⁢           ⁢   D   ⁢           ⁢     sin   ⁡     (       π   ⁢           ⁢   D     +   φ     )           π   ⁢           ⁢   D   ⁢           ⁢   R               (   9   )               
and from (7) and (9) the power supply current is
 
     
       
         
           
             
               
                 
                   
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     The dc component across choke inductor L 1  is zero so, 
     
       
         
           
             
               
                 
                   
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     With I m  and Idd in (5) from (9) and (10) respectively, equation (11) becomes after simple algebraic manipulations,
 
ω C 1π 2   RD =sin π D  sin 2 (π D +ω)(sin π D−πD  cos π D )  (12)
 
     And finally with Idd and I m  from (10) and (9) respectively and with ωC 1  from (12), equation (5), the time (θ) dependent Mosfet drain voltage becomes: 
     
       
         
           
             
               
                 
                   
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     Now, with 
             k   =       R   ⁢           ⁢   1         R   ⁢           ⁢   1     +     R   ⁢           ⁢   2               
the comparator voltage V comp  in  FIG. 1  is:
 
     
       
         
           
             
               
                 
                   
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                   ( 
                   14 
                   ) 
                 
               
             
             
               
                 
                   
                     V 
                     threshold 
                   
                   = 
                   
                     
                       
                         R 
                         ⁢ 
                         
                             
                         
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                         3 
                       
                       
                         
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                           ⁢ 
                           
                               
                           
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                   ( 
                   15 
                   ) 
                 
               
             
           
         
       
     
     It is contemplated here that R 3  can be made variable (digital potentiometer) and thus an adaptive control can be implemented. 
     In reference to  FIG. 1  again—the primary power supply  133  can be a lithium-ion battery pack or another power source. ON/OFF switch  124  is used to turn the instrument on and off. The linear regulator (LDO)  136  provides the necessary low voltage power Vcc at line  125  for the logic circuits and the MCU. Preferably LDO output voltage Vcc is between 3 and 3.6 V. After the instrument is turned on via switch  124  the DC/DC converter  101  is connected to the primary input Power supply  133  via line  123 . Line  122  represents the output voltage of the DC/DC converter, Vdd. C 8  is the output filter capacitor of the converter. The instrument is also provided with a display  119  and an alarm module  120 . The latter can be a buzzer or a regular speaker. For one skilled in the art it is recognized that the power stage consisting of Mosfet transistor Q 1 , RF choke L 1 , capacitor C 1 , C 2 , and inductor L 2  represents a Class E amplifier with a complex load (detailed as blocks  131  or  130 ) coupled directly or via transformer T 1  in node  129 . The load can be piezoelectric, resistive or magnetostrictive. For high power applications a stack consisting of piezoelectric rings can be used. Some manufactures of such rings are: Channel Industries (C8800 series for example), Morgan Matroc or APC. For magnetostrictive loads (terfenol) a typical manufacturer is ETREMA Inc. In this case the load will be mainly inductive and ref [4], [5] provide a way to carry out an inductive impedance inverter for Class E operation. The power stage preferably operates at a frequency near the series resonance of the piezoelectric element (marked fs in the waveform  104  block) where it can be reduced (simplified) to a parallel RC circuit. The closer to resonance frequency fs of the piezoelectric load, the more “pure resistive” the load. The drain voltage in  FIG. 1  represented in block  111  and calculated in formula (13) above can have a shape represented in  FIG. 2 ,  3 ,  4  or  5  depending on the load operating conditions. The operation of a class E amplifier under Class E circuit parameters variation is well known and has been described by Raab[7] and Kazimierczuk [2], [3], however the value of the drain voltage depending on the load phase angle, frequency, power supply voltage Vdd and duty cycle has not been directly calculated. A class E predictor [1] and drain voltage slope [2] have been used as ways to determine the distance from Class E operation. A new, fast feedback method has been implemented in the present description where the reaction time of the feedback loop due to load electrical impedance change is one Power Amplifier clock cycle. Also, a distance from Class E nominal operation is implemented in the form of digital timer content. It is contemplated that for one skilled in the art the present description can be extended to other industrial application some of which are (but not limited to) induction heating and cleaning it is contemplated that although block  102  was described as a “D flip-flop” external to the processor  100 , this could be integrated within the processor or an equivalent circuit could be used (for example an RST flip-flop) without departing from the scope of the present description. Similarly, alternatives to the Mosfet switch Q 1  are within the scope of the current description. Also, although the preferred embodiment presents an “analog” Duty cycle adjustment block  117  using a comparator and an integrator, a “digital” duty cycle adjustment (a Pulse Width Modulator PWM, or a digitally generated variable width pulse) is within the scope of the present description. Similarly, it is contemplated that the Class E DC/DC converter in the preferred embodiment in  FIG. 1  can be replaced with other known DC/DC converter topologies, for example, SEPIC, çuk without departing from the scope of the current description. Linear regulator  136  can also be a step-down (buck) DC/DC converter.