Abstract:
Cycle-by-cycle synchronous waveform shaping is provided for by filtering and combining of square and/or impulse shaped signals. Specifically, a plurality of first square shaped signals is generated and filtered using at least one first filter to produce at least one filtered signal. A plurality of second square shaped signals is generated and filtered using at least one second filter to produce at least one second filtered signal. The at least one first and at least one second filtered signals are combined to produce a continuous shaped waveform having a characteristic shape within each of a plurality of data periods defining a data rate. Alternatively, at least one impulse signal having a plurality of sinusoidal impulses each comprising a positive impulse and a negative impulse is generated. The at least one impulse signal is filtered using at least one filter to produce the continuous shaped waveform.

Description:
CROSS-REFERENCES TO RELATED APPLICATIONS  
       [0001]    This application claims priority from U.S. application Ser. No. 60/301,055, filed Jun. 25, 2001 entitled “Cycle-by-cycle Synchronous Waveform Shaping Circuits Based on Time-domain Superposition and Convolution.” 
     
    
     
       STATEMENT AS TO RIGHTS TO INVENTIONS MADE UNDER FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT  
         [0002]    NOT APPLICABLE  
         REFERENCE TO A “SEQUENCE LISTING,” A TABLE, OR A COMPUTER PROGRAM LISTING APPENDIX SUBMITTED ON A COMPACT DISK.  
         [0003]    NOT APPLICABLE  
         BACKGROUND OF THE INVENTION  
         [0004]    This invention relates generally to techniques for waveform shaping and more specifically to techniques for shaping individual cycles of a carrier waveform.  
           [0005]    Waveform shaping at baseband has been an important process in the transmission of communication signals. Such waveform shaping is generally performed to obtain a more bandwidth efficient signal before modulation onto a carrier that allows transmission over a specific frequency band. Traditional modulation techniques for known modulation schemes such as Frequency Shift Keying (FSK) requires processing multiple cycles of the carrier signal in order for the receiver to lock effectively and detect a single symbol contained in the original signal. Such techniques also generally require the phase of the modulated signal to be continuous. The signal transmitted for a system employing such traditional techniques need not perform waveform shaping on a cycle-by-cycle basis, since symbol is spread over multiple cycles of the carrier waveform. However, when a communication signal represents each symbol using relatively few, or even just one cycle of the carrier waveform, shaping of individual cycles of the carrier waveform becomes necessary. Furthermore, it may still be required that the phase of the modulated signal be continuous.  
         BRIEF SUMMARY OF THE INVENTION  
         [0006]    Cycle-by-cycle synchronous waveform shaping is provided for by filtering and combining of square and/or impulse shaped signals. Specifically, a plurality of first square shaped signals is generated and filtered using at least one first filter to produce at least one filtered signal. A plurality of second square shaped signals is generated and filtered using at least one second filter to produce at least one second filtered signal. The at least one first and at least one second filtered signals are combined to produce a continuous shaped waveform having a characteristic shape within each of a plurality of data periods defining a data rate. In one embodiment, the continuous shaped waveform is a Frequency Shift Keying (FSK) signal having at least a first and a second frequency, wherein the first square shaped signals and the at least one first filter correspond to the first frequency, and wherein the second square shaped signals and the at least one second filter correspond to the second frequency.  
           [0007]    Alternatively, at least one impulse signal having a plurality of sinusoidal impulses each comprising a positive impulse and a negative impulse is generated. The at least one impulse signal is filtered using at least one filter to produce the continuous shaped waveform. In one embodiment, at least one of the sinusoidal impulses is generated by differentially combining a square shaped signal with a delayed version of the square shaped signal. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0008]    [0008]FIG. 1 illustrates a Frequency Shift Keying (FSK) signal that can be generated using a particular technique for cycle-by-cycle synchronous waveform shaping.  
         [0009]    [0009]FIG. 2 illustrates an embodiment of an FSK cycle-by-cycle synchronous waveform shaping circuit in accordance with the present invention.  
         [0010]    [0010]FIG. 3 is a block diagram of an implementation of the FSK cycle-by-cycle synchronous waveform shaping circuit.  
         [0011]    [0011]FIGS. 4A, 4B,  5 A, and  5 B are time domain plots representing the various filtered signals to be differentially combined in order to produce the desired FSK cycle-by-cycle synchronous waveform.  
         [0012]    [0012]FIG. 6 is a time domain plot representing the desired FSK cycle-by-cycle synchronous waveform produced by the implementation shown in FIG. 3.  
         [0013]    [0013]FIG. 7A is a functional diagram of the convolution process used in a second embodiment of the cycle-by-cycle synchronous waveform shaping circuit in accordance with the present invention.  
         [0014]    [0014]FIGS. 7B and 7C illustrate examples of how the convolution process shown in FIG. 7A can be used to generate a Frequency Shift Keying (FSK) or a Binary Phase Shift Keying (BPSK) signal, respectively.  
         [0015]    [0015]FIG. 8 is a block diagram of the second embodiment  800  of the cycle-by-cycle synchronous waveform shaping circuit producing a BPSK signal in accordance with the present invention.  
         [0016]    [0016]FIG. 9 is a time domain plot representing the desired BPSK cycle-by-cycle synchronous waveform produced by the implementation shown in FIG. 8. 
     
    
     DETAILED DESCRIPTION OF THE INVENTION  
       [0017]    [0017]FIG. 1 illustrates a Frequency Shift Keying (FSK) signal that can be generated using a particular technique for cycle-by-cycle synchronous waveform shaping. This technique generates a FSK signal by sending a mixed square waveform through a low pass filter. Within each predefined frame, the mixed square waveform is either a lower frequency square wave or a higher frequency square wave. Thus, the filtered output represents a FSK signal. However, since the mixed square waveform contains both lower and higher frequency square waveforms, the single lowpass filter is not sufficient. This is because the harmonics of the lower frequency square waveforms are not removed. Therefore, the harmonics of the lower frequency square waveforms interfere with the higher frequency components of the output signal. As can be seen in FIG. 1, this approach generates a distorted FSK signal. More effective approaches to cycle-by-cycle synchronous waveform shaping are discussed below.  
         [0018]    [0018]FIG. 2 is a high level functional block diagram of an illustrative embodiment  200  of a FSK cycle-by-cycle synchronous waveform shaping circuit in accordance with the present invention. The circuit  200  produces a FSK cycle-by-cycle synchronous waveform  290  having distinct data periods including data periods  292 ,  294 ,  296 , and  298 . Four synchronous digital signals  201 ,  202 ,  203 , and  204  are provided as inputs to the circuit. The digital signals  201  and  202  each has a cycle of length T during which time the signal level transitions from a high level to a low level, or vice versa. Similarly, the digital signals  203  and  204  each has a cycle of length T/ 2  in which time the signal level transitions from a high level to a low level, or vice versa. Typically, the digital signals  201 ,  202 ,  203 , and  204  can be generated by any of a number of conventional techniques such as digital logic, a processor, or the others implementations.  
         [0019]    The digital signal  201  is passed through a digital block unit  211  and a low pass filter  221 , to produce a filtered signal  231 . The digital signal  202  is passed through a digital block unit  212  and a low pass filter  222 , to produce a filtered signal  232 . The digital signal  203  is passed through a digital block unit  213  and a low pass filter  223 , to produce a filtered signal  233 . Finally, the digital signal  204  is passed through a digital block unit  214  and a low pass filter  224 , to produce a filtered signal  234 . The digital block units  211 ,  212 ,  213 , and  214  each removes the DC component from each of the digital signals  201 ,  202 ,  203 , and  204 , respectively. The filtered signals  231  and  232  combine at a combiner  242  to form a first combined signal  252 . The filtered signals  233  and  234  combine at a combiner  244  to form a second combined signal  254 .  
         [0020]    The first combined signal  252  might include regions in the signal a “null”. Consider for example, the region “A” of the input signals  201 ,  202 . The figure shows that at the region “A”, there is a 180° phase difference between the digital signals  201  and  202 . Consequently, the filtered signals  231  and  232 , which correspond to the digital signals  201  and  202 , significantly cancel each other in the region “A” when they are combined at the combiner  242 . Thus, the first combined signal  252  has an a null signal at a region that corresponds to the region “A”. On the other hand, in the same region of the combined signal  254  that corresponds to region “A”, the signal is amplified. That is, in region “A”, there is a 0° phase difference between the digital signals  203  and  204 . Thus, the filtered signals  233  and  234 , which correspond to the digital signals  203  and  204 , significantly add to each other in the region “A” when they are combined at the combiner  244 .  
         [0021]    Similarly, the second combined signal  254  is effectively a null signal in certain other regions. For example, in an illustrative region  37  B,” there is ideally a 180 degree phase difference between the digital signals  203  and  204 . Consequently, the filtered signals  233  and  234 , which correspond to the digital signals  203  and  204 , significantly cancel each other in the region “B” when they are combined at the combiner  244 . Thus, the second combined signal  254  is effectively a null signal within the region “B.” On the other hand, in the same region, the combined signal  252  is an amplified signal. That is, in region “B,” there is ideally a 0 degree phase difference between the digital signals  201  and  202 . Thus, the filtered signals  231  and  232 , which correspond to the digital signals  201  and  202 , significantly add to each other in the region “B” when they are combined at the combiner  242 .  
         [0022]    The first and second combined signal  252  and  254  are combined to each other at a combiner  260  to form the FSK cycle-by-cycle synchronous waveform  290  suitable for transmission. The waveform  290  has distinct data periods including data periods  292 ,  294 ,  296 , and  298 . Note that data periods  292 ,  294 , and  298  correspond to regions in which the first combined signal  252  contributes a signal having a cycle of length T, and the second combined signal  254  contributes an effectively null signal. Also note that data period  296  corresponds to a region in which the second combined signal  254  contributes a signal having two cycles of length T/ 2  each, and the first combined signal  252  contributes an effectively null signal.  
         [0023]    It can be appreciated from FIG. 2 that the principle of superposition provides an alternate configuration whereby the digital signals  201 - 204  are combined to produce an intermediate digital signal, prior to performing the filtering. The intermediate digital signal can then be DC blocked to remove a DC component if necessary, and then low pass filtered using a single appropriately designed low pass filter.  
         [0024]    [0024]FIG. 3 is a block diagram  300  of an implementation of the FSK cycle-by-cycle synchronous waveform shaping circuit  200 . This implementation produces one cycle of a signal with frequency f 0  (one cycle having a 1/f 0  period) to represent a bit “1” and two cycles of a signal with frequency f 1  (two cycles each having 1/f 1 , period) to represent a bit “0.” Here, f 1 , is a frequency that is twice f 0 .  
         [0025]    A Delayed Lock Loop (DLL) circuit  302  receives a raw data signal  304  and an asynchronous clock signal  306  and performs the function of locking to the timing of the incoming raw data signal  304 . The DLL circuit  302  outputs a Sync Clk signal  308 , a Sync Data signal  310 , and a 2× Sync Clk signal  312 . The Sync Clk signal  308  has a frequency equivalent to the data rate of the Sync Data signal  310 . The 2× Sync Clk signal  312  has a frequency twice the data rate of the Sync Data signal  310 . Both clock signals  308  and  312  are synchronous with the Sync Data signal  310 .  
         [0026]    The Sync Clk signal  308 , Sync Data signal  310 , and 2× Sync Clk signal  312  are input to a Combinational Logic Circuit  314 , which produces a Low Dout signal  321 , a Low Clk signal  322 , a High Dout signal  323 , and a High Clk signal  324 . The Low Dout signal  321  passes through a coupling capacitor  331  and a low pass filter  341  to form a filtered signal  351 . The Low Clk signal  322  passes through a coupling capacitor  332  and a low pass filter  342  to form a filtered signal  352 . The High Dout signal  323  passes through a delay block  326 , a coupling capacitor  333 , and a low pass filter  343  to form a filtered signal  353 . The High Clk signal  324  passes through a delay block  328 , a coupling capacitor  334 , and a low pass filter  344  to form a filtered signal  354 .  
         [0027]    Note that the Low Dout signal  321  and the Low Clk signal  322  together represent cycles of the lower frequency f 0  signal used to indicate the bit “1”s. However, in this implementation, the Low Dout signal  321  alone carries the information relating to the location of the bit “1”s. The Low Clk signal  322  is merely a clock signal synchronous with the Low Dout signal  321 . Nevertheless, the Low Clk signal  322  is used in combination with the Low Dout signal  321  to ensure that the time span of a non-zero value on either digital signal  321  or  322  will be at most 2T L , where T L  is the time span between two possible transitions on either signal  321  or  322 .  
         [0028]    Similarly, the High Dout signal  323  and the High Clk signal  324  together represent cycles of the higher frequency f 1  signal used to indicate the bit “0”s. The High Dout signal  323  alone carries the information relating to the location of the bit “0”s. The High Clk signal  324  is merely a clock signal synchronous with the High Dout signal  323 . The two signals used in combination ensure that the time span of a non-zero value on either digital signal  323  or  324  will be at most 2T H , where T H  is the time span between two possible transitions on either signal  323  or  324 .  
         [0029]    Also note that the low pass filters  341  and  342  together form a low pass filter group  1  in which each filter has a cut-off frequency corresponding to the pulse frequency ½T L  of the digital signals (Low Dout signal  321  and Low Clk signal  322 ) they serve. The low pass filters  343  and  344  together form a low pass filter group  2  in which each filter has a cut-off frequency corresponding to the pulse frequency ½T H  of the digital signals (High Dout signal  323  and High Clk signal  324 ) they serve. The low pass filters  321 ,  322 ,  323 , and  324  thus appropriately reduce the harmonics in the various signals being filtered. The low pass filters  321 , 322 ,  323 , and  324  can be implemented as analog infinite response impulse response filters. Any kind of appropriate conventionally known filter can be used, including Butterworth filters, Bessel filters, and so on. In a particular embodiment of the invention, for example, the low pass filters are implemented as Gaussian filters, which are known to contribute less distortion in neighboring pulses of the signals being filtered.  
         [0030]    Delay blocks  326  and  328  are used to add delay to the High Dout signal  323  and High Clk signal  324  in order to compensate for the difference between the delay associated with low pass filter group  1  and the delay associated with low pass filter group  2 . The delay blocks  326  and  328  can be implemented as adjustable digital delays, a long transmission path or wire, or others.  
         [0031]    Referring again to FIG. 3, the filtered signals  351  and  352  are differentially combined at a differential combiner  360  to produce a first differentially combined signal  364 . Within each region representing a data period associated with a bit “0,” the filtered signals  351  and  352  significantly cancel each other at the differential combiner  360 , and the first differentially combined signal  364  is effectively a null signal within the region. Similarly, the filtered signals  353  and  354  are differentially combined at a differential combiner  362  to produce a second differentially combined signal  368 . Within each region representing a data period associated with a bit “1,” the filtered signals  353  and  354  significantly cancel each other at the differential combiner  362 , and the second differentially combined signal  368  is effectively a null signal within that region.  
         [0032]    The first and second differentially combined signals  364  and  368  are differentially combined to each other at a differential combiner  370  to produce the desired FSK cycle-by-cycle synchronous waveform  290  that is suitable for transmission. Note that differential combiners  360 ,  362 , and  370  are used because the various signals are transmitted in a differential mode, which allows improvements in noise rejection and formation of sinusoidal waveforms. Differential signaling in this embodiment is achieved by using the combinatorial logic circuits  314  to appropriately control the polarity of the Low Dout signal  321 , the Low Clk signal  322 , the High Dout signal  323 , and the High Clk signal  324 .  
         [0033]    It should be noted that while FIG. 3 illustrates the production of an FSK cycle-by-cycle synchronous waveform, a similar implementation can be used to generate a Binary Phase Shift Keying (BPSK) or another type of Phase Shift Keying (PSK) cycle-by-cycle synchronous waveform by generating digital signals of different phases and filtering and/or combining such digital signals.  
         [0034]    [0034]FIGS. 4A, 4B,  5 A, and  5 B are time domain plots representing the various filtered signals to be differentially combined in order to produce the desired FSK cycle-by-cycle synchronous waveform  290 . FIG. 4A and 4B represent the filtered signals  351  and  352 , respectively. Note that these two signals are characterized by the time span T L . FIGS. 5A and 5B represent the filtered signals  353  and  354 , respectively. Note that these two signals are characterized by the time span T H . FIG. 6 is a time domain plot representing the desired FSK cycle-by-cycle synchronous waveform  290  produced by the circuit shown in FIG. 3.  
         [0035]    [0035]FIG. 7A is a functional diagram of the convolution process used in a second embodiment  800  (FIG. 8) of the cycle-by-cycle synchronous waveform shaping circuit in accordance with the present invention. A data pulse  702  and a delayed data pulse  704  are differentially combined at a differential combiner  706  to produce an impulse pair  710  having a positive impulse  712  and a negative impulse  714 .  
         [0036]    The delayed data pulse  704  is delayed in time by a precise amount relative to the data pulse  702  but otherwise resembles the data pulse  702 . The data pulse  702  and delayed data pulse  704  can be generated by digital logic, a processor, or the others implementations. The data pulse  702  and the delayed data pulse  704  overlap in a period of length T/ 2 −Ts. When differentially combined, the data pulse  702  and the delayed data pulse  704  cancel each other in this overlapping period, and non-overlapping portions of the pulses  702  and  704  form a positive impulse  712  and a negative impulses  714  of an impulse pair  710 .  
         [0037]    The impulse pair  710  is convolved with a Gaussian filter  720  in the time domain to produce a sinusoidal pulse  730  having a positive half cycle  732  and a negative half cycle  734 . The positive impulse  712  of the impulse pair  710  produces the positive half cycle  732 , which resembles the impulse response of the Gaussian filter  720 . The negative impulse  714  of the impulse pair  710  produces the negative half cycle  734 , which resembles the negative of the impulse response of the Gaussian filter  720 . The Gaussian filter  720  has a compact impulse response and a less oscillatory nature compared to other filter designs. The Gaussian filter  720  can also be realized in the form of a LC circuit. However, other types of filters such as Butterworth filters and Bessel filters may also be used.  
         [0038]    [0038]FIGS. 7B and 7C illustrate examples of how the convolution process shown in FIG. 7A can be used to generate a Frequency Shift Keying (FSK) or a Binary Phase Shift Keying (BPSK) signal, respectively. The convolution process shown in FIG. 7A is highly controllable and precise in generating a sinusoidal pulse at a specified time. By generating and superpositioning appropriate sinusoidal pulses at particular positions in time, appropriate data modulated signals such as FSK and BPSK signals can be produced. FIG. 7B illustrates that a portion of an FSK signal can be produced by concatenating a sinusoidal impulse having a length of  2 T with two sinusoidal impulses each having a length of T. FIG. 7C illustrates that a portion of a BPSK signal can be produced by concatenating a sinusoidal impulse having a length of T with another sinusoidal impulse having a length of T but being inverse in amplitude.  
         [0039]    [0039]FIG. 8 is a block diagram of the second embodiment  800  of the cycle-by-cycle synchronous waveform shaping circuit producing a BPSK signal in accordance with the present invention. Here, two distinct sinusoidal pulses  802  and  804  are generated at particular positions in time and differentially combined to form one portion of a desired BPSK cycle-by-cycle synchronous waveform  806 . Although only the sinusoidal pulses  802  and  804  are shown in FIG. 8, it should be understood that other sinusoidal pulses preceding, following, or even overlapping with sinusoidal pulses  802  and  804  are also differentially combined to form other portions of the BPSK cycle-by-cycle synchronous waveform  806 .  
         [0040]    Referring to FIG. 8, a digital signal  810  containing data pulses of length T is generated and provided to the circuit  800 . An AND function block  811  receives the digital signal  810  and a clock signal  812 , which has pulses of length T/ 2  and is synchronous with the digital signal  810 . The AND function block  811  outputs a half-cycle signal  813 . In this manner, each data pulse in digital signal  810  representing a bit ‘1’ (or bit ‘high’) is extracted and reduced to half duty cycle, producing the half-cycle signal  813 . A delay block  814  receives the half-cycle signal  813 , introduces a delay of T s , and produces a delayed half-cycle signal  815 . The half-cycle signal  813  and the delayed half-cycle signal  815  are differentially combined at a differential combiner  816  to produce an impulse pair signal  818 .  
         [0041]    The digital signal  810  is inverted at an inverter  819 , producing an inverted digital signal  820 . An AND function block  821  receives the inverted digital signal  820  and the clock signal  812 , which has pulses of length T/ 2  and is synchronous with the inverted digital signal  820 . The AND function block  811  outputs a half-cycle signal  823 . In this manner, each data pulse in digital signal  810  representing a bit ‘0’ (or bit ‘low’) is extracted and reduced to half duty cycle, producing the half-cycle signal  823 . A delay block  824  receives the half-cycle signal  823  , introduces a delay of T s , and produces a delayed half-cycle signal  825 . The half-cycle signal  823  and the delayed half-cycle signal  825  are differentially combined at a differential combiner  826  to produce an impulse pair signal  828 .  
         [0042]    An impulse regenerating circuit  830  receives the impulse pair signal  818  and produces a regenerated impulse pair signal  832 . Similarly, an impulse regenerating circuit  840  receives the impulse pair signal  828  and produces a regenerated impulse pair signal  842 . Under certain conditions, the impulse pair signals  818  and  828  may not have proper signal level and/or form to be adequate impulse signals. For example, a low slew rate associated with the digital signals  813 ,  815 ,  823 , and  825  caused by digital data buffers supplying these signals may result in a “smearing” of the positive pulses and negative pulses of the impulse pair signals  818  and  828 . These positive and negative pulses could thus lack proper signal level and/or form. The impulse regenerating circuits  830  and  840  corrects such problems by adjusting the signal levels and/or other characteristics of the regenerated impulse pair signals  832  and  842  such that they provide adequate impulse signals.  
         [0043]    A differential combiner  854  receives the regenerated impulse pair signals  832  and  842  and produces a combined regenerated impulse pair signal  852 . A Gaussian filter  854  of length T/ 2 −T s  receives the combined regenerated impulse pair signal  852  and produces the BPSK cycle-by-cycle synchronous waveform  806 . Alternatively, the regenerated impulse pair signal  832  and the regenerated impulse pair signal  842  can be separately filtered and then differentially combined. In such case, two Gaussian filter are needed. FIG. 9 is a time domain plot representing the desired BPSK cycle-by-cycle synchronous waveform produced by the implementation shown in FIG. 8.  
         [0044]    It should be noted that while FIG. 8 illustrates the production of a BPSK cycle-by-cycle synchronous waveform, a similar implementation can be used to generate an FSK cycle-by-cycle synchronous waveform by generating impulse pairs corresponding to different frequencies and filtering and/or combining such impulse pairs.  
         [0045]    Although the present invention has been described in terms of specific embodiments, it should be apparent to those skilled in the art that the scope of the present invention is not limited to the described specific embodiments.  
         [0046]    The specification and drawings are, accordingly, to be regarded in an illustrative rather than a restrictive sense. It will, however, be evident that additions, subtractions, substitutions, and other modifications may be made without departing from the broader spirit and scope of the invention as set forth in the claims.