Abstract:
An apparatus comprising an input stage, an output stage, a bias circuit and a feedback circuit. The input stage may be configured to generate a plurality of intermediate signals in response to an input signal. The output stage may be (i) DC coupled to the input stage and (ii) configured to generate an output signal in response to the intermediate signals. The output stage generally comprises a plurality of distributed amplifiers each configured to receive one of the intermediate signals. The bias circuit may be (i) connected between the input stage and the output stage and (ii) configured to adjust an input impedance of the input stage.

Description:
FIELD OF THE INVENTION 
     The present invention relates to a method and/or architecture for implementing amplifiers generally and, more particularly, to a method and/or architecture for implementing a direct coupled distributed amplifier. 
     BACKGROUND OF THE INVENTION 
     Direct coupled amplifiers operating beyond 50 GHz are needed for test instrumentation, fiber optic systems, and satellite communication systems. Conventional lumped element analog feedback topologies can easily obtain direct coupled performance, but are typically limited in bandwidth by the RC parasitics of the transistors and interconnects. Conventional distributed amplifier topologies achieve wide bandwidths by absorbing transistor and interconnect parasitics into the design, but are difficult to directly DC couple. Conventional distributed amplifier topologies also tend to implement large termination bypass capacitors and are often spatially inefficient. 
     Several conventional approaches are presently used to satisfy the need for high gain-bandwidth direct-coupled amplification. One approach is a resistive feedback lumped element analog topology, such as the Darlington feedback amplifier. However, the performance of a Darlington feedback amplifier ultimately impaired by device and interconnect parasitics. Another approach used is to directly cascade distributed amplifiers using source/emitter follower and/or diode level shift circuits to obtain higher gain-bandwidth. However, directly cascaded conventional distributed amplifiers suffer from the expense of size, stability, poor inter-stage voltage standing wave ratio (VSWR) and/or group delay and/or gain ripple. 
     Referring to FIG. 1, a conventional Darlington amplifier  10  is shown. The conventional Darlington amplifier  10  is noted for having wide bandwidth capability. Because of the direct-coupled topology, the conventional Darlington amplifier  10  allows gain performance down to DC. The upper bandwidth performance is, however, ultimately limited by device and interconnect parasitics which have a profound impact at microwave frequencies. 
     In order to achieve high gain-bandwidth, distributed amplifiers have been directly coupled through the use of emitter or source followers and/or diode level shifter circuits. FIG. 2 illustrates a conventional emitter or source follower (or circuit)  20 . FIG. 3 illustrates a conventional diode level shifter (or circuit)  30 . The circuit  20  and the circuit  30  provide DC coupling between distributed amplifiers at the expense of higher power consumption, larger size (about twice the die area of a single stage distributed amplifier) and potential instability and gain ripple problems. In addition, conventional distributed amplifiers incorporate large termination bypass capacitors (not shown) to obtain base band performance and thus implement several off-chip components. 
     Referring to FIG. 4, a performance of a traditional analog direct coupled feedback amplifier is shown. Analog direct coupled feedback amplifiers are characteristic of flat responses down to baseband because of the absence of frequency limiting capacitor and inductor networks. A forward transmission (i.e., insertion) scattering parameter (i.e., S 21 ) can have a clean performance from baseband to the 3-db BW requirement of greater than 35 GHz for a 40 Gb/s operation. An input reflection scattering parameter (i.e., S 11 ) should also be consistent over the bandwidth. In particular, the parameter S 11  should reflect a constant impedance across the band to ensure flat transimpedance performance. The output reflection scattering parameter (i.e., S 22 ) should be better than 10 dB across the band. The log plot shows that the direct coupled topology achieves consistent baseband behavior. 
     Referring to FIG. 5, a linear frequency performance of the traditional analog direct coupled topology is shown. Gain and input return-loss degrade at higher frequencies due to device and interconnect parasitics. Specifically, capacitive parasitics tend to roll-off the forward transmission gain (i.e., S 21 ) as the frequency increases. The input return-loss (i.e., input reflection coefficient S 11 ) reduces the input impedance at higher frequencies. The output reflection coefficient (i.e., S 22 ) remains relatively constant at all frequencies. 
     It would be desirable to obtain the consistent baseband performance of a direct coupled device while being implemented in a compact area while achieving high frequency bandwidth performance similar to performance obtained from distributed amplifier designs. 
     SUMMARY OF THE INVENTION 
     The present invention concerns an apparatus comprising an input stage, an output stage, a bias circuit and a feedback circuit. The input stage may be configured to generate a plurality of intermediate signals in response to an input signal. The output stage may be (i) DC coupled to the input stage and (ii) configured to generate an output signal in response to the intermediate signals. The output stage generally comprises a plurality of distributed amplifiers each configured to receive one of the intermediate signals. The bias circuit may be (i) connected between the input stage and the output stage and (ii) configured to adjust an input impedance of the input stage. 
     The objects, features and advantages of the present invention include providing a direct coupled distributed amplifier that may (i) have a distributed common emitter input stage directly DC coupled to a second common emitter, (ii) implement a Darlington distributed stage that is scalable, (iii) provide cascadable wide band performance, (iv) have scalability to N number of directly DC coupled distributed amplifier sections for achieving higher direct-coupled gain, (v) have a self-biasing portion that incorporates broadband active load, (vi) have termination that enhances and maintains lower frequency response, (vii) have a biasing portion that may include feedback between the active components of distributed stages, (viii) have a biasing portion that provides gain-temperature compensation, (ix) implement an application of the Darlington pair as a transconductor cell of a microwave distributed amplifier topology, and/or (x) be an application of direct-coupled feedback between 2 or more stages of gain within a distributed amplifier. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     These and other objects, features and advantages of the present invention will be apparent from the following detailed description and the appended claims and drawings in which: 
     FIG. 1 is a diagram illustrating a conventional Darlington amplifier; 
     FIG. 2 is a diagram illustrating conventional distributed amplifiers being directly cascaded by an emitter follower; 
     FIG. 3 is a diagram illustrating conventional distributed amplifiers being directly cascaded by diodes; 
     FIG. 4 is a graph illustrating a traditional analog feedback amplifier multi-decade response; 
     FIG. 5 is a graph illustrating a traditional analog feedback amplifier high frequency response; 
     FIG. 6 is a circuit diagram illustrating a direct coupled distributed amplifier in accordance with a preferred embodiment of the present invention; 
     FIG. 7 is a diagram illustrating an example for extending the number of amplifier cells; 
     FIG. 8 is a diagram illustrating another example for extending the number of amplifier cells; 
     FIG. 9 is an example of an alternate amplifier cell; 
     FIG. 10 is a graph illustrating a multi-decade response of the present invention; 
     FIG. 11 is a graph illustrating a direct, coupled distributed amplifier high frequency response; 
     FIG. 12 is a graph illustrating a direct coupled distributed amplifier S-parameter temperature dependence; and 
     FIG. 13 is a graph illustrating a self bias compensating for gain over temperature where ICC is nearly proportional to T(K). 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     The present invention applies the distributed amplification technique that may enhance a traditional feedback amplifier to obtain higher gain and bandwidth per unit area and power consumption than a traditional distributed amplifier. The present invention may directly (DC) couple a number of distributed amplifier stages in order to increase gain and/or bandwidth (BW) without compromising low frequency performance. Another feature of the present invention implements a biasing circuit to provide active load terminations in order to extend the low frequency response while using fewer off-chip components. 
     Referring to FIG. 6, a circuit  100  is shown implementing a preferred embodiment of the present invention. The circuit  100  illustrates a two section distributed 2-stage cascaded amplifier. The circuit  100  generally comprises an input stage  102 , an output (or amplifier) stage  104 , and a bias stage  106 . 
     The input stage  102  may be implemented as a distributed common-emitter amplifier. The input stage  102  generally comprises a transistor Q 11  and a transistor Q 12 . The transistors Q 11  and Q 12  are generally directly DC coupled to the output stage  104 . The input stage  102  may be modified to implement Darlington pairs (not shown), similar to the output stage  104 . 
     The output stage  104  generally comprises a number of distributed Darlington amplifier stages  110   a - 110   n . The output stage  104  may be implemented as a distributed direct-coupled amplifier. The stage  110   a  generally comprises a transistor Q 21   d , a transistor Q 21 , and a bias resistor R 21 . The stage  110   n  generally comprises a transistor Q 22   d , a transistor Q 22 , and a bias resistor R 22 . While a distributed Darlington amplifier has been described as an example implementation of the second stage  104 , other implementations, such as a common-emitter, cascode, emitter-follower/common-emitter or other device configuration using a combination of FET and BJT technologies may be implemented to meet the design criteria of a particular implementation. 
     An intermediate broadband transmission line (TLIN_LC 1 ) is generally shared between the input stage  102  and the output stage  104 . The transmission line TLIN_LC 1  may be optimized to other than a 50 ohm system impedance in order to enhance RF performance. The transmission line TLIN_LC 1  may be implemented as a high impedance, high inductance transmission line. 
     The bias stage  106  may be implemented as a self bias portion that may (i) regulate the bias current of both the input stage  102  and the amplifier stage  104 , (ii) provides a broadband low impedance termination for an output transmission line (e.g., TLIN_OUT) and intermediate synthetic transmission line TLIN_LC 1 , (iii) employs a feedback resistor which provides DC stabilization and/or enhance RF performance, (iv) provides gain-temperature compensation. The bias stage  106  may be implemented as an active bias circuit. 
     Further, the input stage  102  and the amplifier stage  104  may be scaled to include more sections along a synthetic input, the intermediate, and the output transmission lines, TLIN_IN, TLIN_LC 1 , and TLIN_OUT, respectively (to be described in more detail in connection with FIGS.  7  and  8 ). The amplifier  104  may be scaled to include more direct-coupled cascaded stages along the cascaded transmission line path TLIN_B, TLIN_C, TLIN_BB, TLIN_CC, effectively directly cascading more distributed stages. 
     The transmission lines TLIN_IN and TLIN_B generally construct an artificial (or synthetic) transmission line with the capacitive input impedance of the common-emitter transistors Q 11  and Q 12 . The input transmission line TLIN_IN is terminated by an R-C network comprising a resistor RTERM_IN and a capacitor CBYPIN. An intermediate synthetic transmission line TLIN_LC 1  generally comprises a line TLIN_C 1 , an output feed line TLIN_C of the common-emitter stage  102  and input feed line TLIN_BB. The synthetic transmission line TLIN_LC 1  generally works in combination with the effective shunt capacitance provided by the output impedance of the transistors Q 11  and Q 12  and input impedance of the Darlington amplifier cells  110   a - 110   n . The outputs of the common emitter stage  102  is directly-coupled to the input of the output stage  104 , thus preserving baseband performance. The intermediate transmission line TLIN_LC 1  is terminated on one side by an RC termination comprising the resistor RTERM_N 2  and the capacitor CBYP 2 , and on the other side by the resistor RTERM_N 1  and the capacitor CBYP 1 . In addition, the active bias network  106  may also provide a controlled low impedance at a port  112  in parallel with the capacitor CBYP 1  and series with the resistor RTERM_N 1 . The active load generally extends the low frequency response by providing a low frequency bypass impedance. The low frequency impedance is approximately equal to Rb 2 +1/gmb 1 +(1/gmb 2 +Rb 2   e )/Beta+(Rb 1 /Beta 2 )+1/sCb 2 *Beta 2 , effectively extending the bypass capacitance CBYP 1  to (Cbyp+Cb 2 *Beta 2 ). 
     As discussed, the distributed output stage  104  generally comprises a plurality of Darlington Amplifier cells  110   a - 110   n . In the illustration shown in FIG. 6, the Darlington cell  110   a  generally comprises a transistor Q 21   d  and a transistor Q 21  and a bias resistor R 21 . The Darlington cell  100   n  generally comprises a transistor Q 22   d  and a transistor Q 22  and a bias resistor R 22 . The Darlington cells  110   a - 110   n  may be used in place of common emitter transistors. In general, the Darlington cells  110   a - 110   n  have higher input impedance and lower effective input capacitances, lower input loss, and/or ability to provide higher output current and voltage drive levels when compared with other cells. In addition, the Darlington cells  110   a - 110   n  provide approximately two diode level shifts which sets the collector to emitter voltage VCE on the common emitter transistors Q 11  and Q 12 . The level shift enables the direct DC coupling of an RF/DC feedback resistor RFB and transmission line (e.g., Tlin-fb) from the emitters of the transistors Q 21   d  and Q 22   d  to the bases of the transistor Q 11  and Q 22 . The DC coupling may set up a DC self-biasing loop in addition to providing control of the transimpedance, gain, and broadband input impedance of the amplifier. While Darlington cells may be preferred, a common-emitter/common-emitter, common-collector/common-emitter, common-emitter/common-source, or common-source/common-emitter cell configuration may be used while achieving direct-coupled feedback from the output stage to the input stage. 
     The output synthetic transmission line TLIN_OUT generally is synthesized by the transmission lines TLIN_OUT and TLIN_CC in conjunction with the effective output capacitance impedances of the Darlington amplifier cells  110   a - 110   n . An output transmission line R-C termination generally comprises a resistor RTERM_OUT and a capacitor CBYP 3 . The active bias circuit  106  is configured and applied in parallel to the termination bypass capacitor CBYP 3  and provides both an apparatus for self-biasing the direct-coupled distributed amplifier  104  with a single supply voltage (e.g., VCC). The bias circuit  106  also provides a controlled low impedance at a port  114  in parallel with the termination bypass capacitor CBYP 3  in order to extend the low frequency performance of the amplifier  100 . The effective impedance at port  114  is expressed by the following equation: 
     
       
           Rb   2   e +1/gmb 2 +( Rb   1 /Beta 1 )∥(1 /sCb   2 *Beta 1 ). 
       
     
     This increases the effective bypass capacitance from the capacitance CBYP to (CBYP 3 +CBYP 2 *Beta), since there is effectively a shunt capacitor of CBYP 2 *Beta in parallel with CBYP 3 . 
     The direct coupled distributed amplifier  100  may be extended to N number of multiple sections along the synthetic transmission lines TLIN_IN, TLIN_LC 1  and TLIN_OUT. The direct-coupled distributed amplifier  100  may be extended to M number of cascaded stages (e.g., from input to output) in order to increase the direct-coupled gain performance. Furthermore, the Darlington amplifier cells  110   a - 110   n  may be replaced by an emitter-follower/common-source or common-source/common-emitter transistor configuration without departing from the spirit of the direct-coupled feedback nature of the invention. 
     The biasing scheme may be expressed by the following approximate equations: 
     
       
           Ic   1 = Ic   11 = Ic   12 =( Vcc −4 *Vbe−Ic   2 * Rb   2   e )/[2*( Rb   1 + Rb   2 + R term —   n   1 )]  1. 
       
     
     
       
           Ic   2 = Ic   22 = Ic   21 = Ic   1 *( Ae   2 / Ae   1 ) where  Ae   1 =the area of  Q   11  and  Q   12 ,  Ae   2 =the area of  Q   21  and  Q   22 )  2. 
       
     
     
       
           Ic   2   d=Ic   22   d=Ic   21   d=Vbe/R   22 = Vbe/R   21   3. 
       
     
     
       
           Vce   1 = Vce   11 = Vce   12 = 2   Vbe   4. 
       
     
       Vce _darlington= Vcc −2 *Ic   1 * Rb   1 − Vbe −2*( Ic   2 + Ic   2   d )*( Rb   2   e+R term_out)  5. 
     Temperature dependence 
     
       
         δ Ic   2   d/dT =(δ Vbe/δT )*(1 /R   22 )  1. 
       
     
     
       
         δ Ic   1 /δ T=δIc   2 /δ T =(δ Vbe/δT )*[4−(2 /Rb   2   e )*(δ Ic   1 /δ T )]/[2*( Rb   1 + Rb   2 + R term —   n   1 )]  2. 
       
     
     where (δVbe/δT)=−1 to −2 mV/C depending on the technology. 
     Note these equations are approximate and will depart from the ideal case when DC beta becomes extremely low (&lt;10) or RFB becomes significantly high (&gt;500 ohms). 
     The direct coupled distributed amplifier  100  was implemented using commercially available InP heterostructure bipolar transistor (HBT) technology with fT=175 GHz and Fmax=200 GHz. However, the present invention may be applied to other semiconductor technologies such as SiGe HBT, GaAs HBT, pseudomorphic high-electron mobility transistor (PHEMT), Bi-CMOS and other appropriate technologies. 
     The direct coupled distributed amplifier  100  may be particularly useful for a 43 Gb/s return-to-zero photo-receiver application. The bandwidth needed is greater than 43 GHz with a target of 50 GHz. Simulation of this design are shown compared to a traditional resistive feedback pre-amplifier transimpedance design. The target small signal gain and bandwidth is 10 dB and 50 GHz, respectively. Direct-coupled baseband gain performance below 1 MHz was also a target specification. 
     The new DC coupled distributed amplifier  100  can be useful in many applications inclusive of, but not limited to (i) Mach-Zenhder optical modulator drivers, (ii) Eletro-absorption modulator drivers, (iii) Transimpedance amplifiers, (iv) Wide-band switch architectures, (v) Wideband 2-18 GHz electronic warfare (EW), (vi) Wideband test equipment, and/or (vii) Wideband Bi-CMOS amplifiers for 10 and 40 Gb/s fiber applications as well as 10 GHz microprocessor applications. 
     The present invention may be used to satisfy a 40 Gb/s Return-to-Zero transimpedance preamplifier application which may need bandwidths in excess of 50 GHz. The product may be implemented using InP single heterojunction bipolar transistor (SHBT) process technology available from Global Communication Semiconductors, Inc. (GCS). However, other technologies may be used. This circuit  100  can also be offered as a wide-band instrumentation amplifier for test equipment. Reconfigured, the circuit  100  may also be applied to wideband military EW radar and communication systems and/or fiber optic transmitter systems where high broadband power is needed. The circuit  100  may have applicability to microwave switch buffer amplification (e.g., single pole double throw (SPDT), single pole quad throw (SP 4 T), etc.) or active switch applications where baseband performance down to 10 KHz is needed. 
     Referring to FIG. 7, an example of a circuit  100 ′ is shown illustrating a plurality of elements  110   a - 110   n . A number of amplifier cells  110   a - 110   n  are shown in the horizontal direction. A corresponding number of input cells (e.g., the transistors Q 11  and Q 12 ) are also shown. 
     Referring to FIG. 8, an example of a circuit  100 ″ is shown illustrating a plurality of elements  110   a ′- 110   n ′. A number of amplifier cells  110   a - 110   a ′ and  110   n - 110   n ′ are shown implemented in the vertical direction. The number of input cells (e.g., the transistors Q 11  and Q 12 ) generally match the number of columns of cells (e.g.,  110   a - 110   n ′ and  110   n - 110   n ′). In general, the number of vertical cells and the number of horizontal cells may be increased or decreased to meet the design criteria of a particular implementation. 
     Referring to FIG. 9, an example of alternate amplifier elements  110   a ′,  111   a ′,  112   a ′ is shown implemented with CMOS Field Effect Transistors (FETs). The amplifier cells  110   a - 110   n  of the output stage and/or the transistors Q 11  and Q 12  of the input stage  102  may be implemented using bipolar transistors FETs, or other transistor types. 
     The following circuit simulations of the invention are based on a commercially available InP HBT technology with fT=150 GHz and fmax&gt;200 GHz. The simulations were executed using ADS software by Agilent. The target application is a 43 Gb/s Return-to-Zero optical receiver which requires a baseband to 50 GHz transimpedance pre-amplifier. The invention is compared with a traditional pre-amplifier simulated in the same technology. 
     FIG. 10 is a graph illustrating a multi-decade response of the present invention. Baseband, and high 3 dB bandwidth is achieved by employing the circuit  100 . Simulation of the forward transmission scatter parameter (e.g., S 21 ) predicts a gain of approximately 10 dB beyond 43 GHz. The input reflection scatter parameter (erg., S 11 ) generally remains well below −10 dB beyond 43 GHz. The output reflection scatter parameter (e.g., S 22 ) also remains below −10 dB below approximately 52 GHz. 
     FIG. 11 is a graph illustrating high frequency response of the present invention. Bandwidth is generally enhanced by a factor of two when compared with the conventional approach of FIG. 5. A flat input impedance (e.g., as indicated by parameter S 11 ) is generally achieved over the frequency band which helps produce a flatter transimpedance bandwidth response. In contrast, conventional distributed amplifiers will appear to have periodic peaks and valleys in the input impedance. Typically the peaks in conventional amplifiers can be as high as −10 dB return loss and valleys can be as low as −30 dB return-loss. The difference in these peaks and valleys are due to the periodic change in input impedance which may create ripple in the transimpedance bandwidth response. The topology of this invention, especially the featured feedback resistance RFB, allows control of the impedance (e.g., obtaining a flatter input impedance). 
     FIG. 12 is a graph illustrating temperature dependent S-parameter of the present invention. The self bias circuit  106  provides first order gain compensation (e.g., &lt;0.25 dB at DC and &lt;2 dB at 50 GHz) over temperature and frequency (e.g., as indicated by the forward transmission gain S 21  and the input reflection coefficient S 11 ). Low gain temperature compensation is achieved by employing a first order proportional to absolute temperature (PTAT) bias scheme which compensates for transconductance temperature dependence. 
     FIG. 13 is a graph illustrating the results of bias circuit  106  compensating for gain over temperature with ICC being nearly proportional to T(K). The bias circuit  106  operates nearly PTAT to provide first order gain-temperature compensation. At low frequencies (e.g., around 1 GHz), the temperature dependence of the device transconductance (e.g., gm) determines the temperature dependence of low frequency gain. A PTAT type of biasing circuit  106  helps insure constant gm versus temperature. At 1 GHz, the forward transmission gain S 21  is maintained to within 0.2 dB over the 0 C to 125 C temperature range. At higher frequencies (e.g., 50 GHz), the forward transmission gain S 21  change is due to the other device parameter variations with temperature. The main parameters affecting the high frequency gain (in addition to the device transconductance gm) are Ccb, Rb, and fT. Still, with the PTAT biasing scheme, the high frequency forward transmission gain S 21  (e.g., 50 GHz) degrades by only by 1.5 dB over the 0 C to 125 C temperature range. 
     The transistors described herein may be implemented as bipolar junction transistors (BJTs), heterojunction bipolar transistors (HBTS) or BiCMOS transistors. However, other transistors with similar characteristics may be implemented to meet the design criteria of a particular implementation. In particular, the various transistors of the present invention may be implemented using a variety of process technologies. For example, any or all of Silicon Germanium (SiGe), Indium Gallium Phosphorous (InGaP), Indium Phosphide (InP), or Gallium Arsenide (GaAs) may be used. However, other process technologies may be implemented to meet the design criteria of a particular implementation. 
     While the invention has been particularly shown and described with reference to the preferred embodiments thereof, it will be understood by those skilled in the art that various changes in form and details may be made without departing from the spirit and scope of the invention.