Abstract:
A data transmission apparatus of an orthogonal frequency multiplex modulation system wherein data is transmitted using a plurality of carriers which are in a mutually orthogonal relationship with one another. The apparatus includes a transmitter which previously inserts a group of predetermined synchronization symbols into an OFDM modulated transmission signal every fixed period and a receiver which demodulates a received OFDM modulated transmission signal to a baseband OFDM signal. In the receiver, an absolute value of an A/D converted digital signal is taken and the signal is bandwidth-limited to a predetermined band width. Then, it is decided whether an amplitude of the signal is larger than or smaller than a predetermined value and a decision result thereof is produced. A null section in the synchronization symbol group is detected on the basis of the decision result and a start point of another synchronization symbol subsequent to the null section is further detected. The time when both of the presence of the null section and the start point of another synchronization symbol are detected is regarded as a synchronization timing and the operation timing of a demodulator for the receiver is set to the synchronization timing.

Description:
CROSS-REFERENCES TO RELATED APPLICATIONS 
     U.S. Pat. application Ser. No. 09/099,390 filed on Jun. 18, 1998 in the names of Atsushi Miyashita et al., entitled IIOFDM MODULATOR AND OFDM MODULATION METHOD FOR DIGITAL MODULATED WAVE HAVING GUARD INTERNAL”, now U.S. Pat. No. 6,304,611, and claiming priority based on Japanese Patent Application No. 9-162579 filed on Jun. 19, 1997 and assigned to the assignee of the present invention is related to the present invention and the disclosure thereof is hereby incorporated by reference. Further, U.S. patent application Ser. No. 09/098,346, filed on Jun. 17, 1998 in the name of Toshiyuki Akiyama et al., entitled “TRANSMITTING AND RECEIVING METHOD OF ORTHOGONAL FREQUENCY DIVISION MULTIPLEXED MODULATION -SIGNAL AND COMMUNICATION SYSTEM”, now abandoned, and claiming priority based on Japanese Patent Application No. 9-161486 filed on Jun. 18, 1997 and assigned to the assignee of the present invention is also related to the present invention and the disclosure thereof is hereby incorporated by reference. 
    
    
     BACKGROUND OF THE INVENTION 
     The present invention relates to a transmission apparatus for transmitting information by means of an Orthogonal Frequency Division Multiplex (OFDM) system and more particularly to the technique for improving the accuracy of detection of synchronization of a demodulator in a receiver. 
     In the OFDM system, a multiplicity of carriers having phases orthogonal to one another and having a narrow frequency band width are arranged in a given frequency band for communication. When a signal modulated by the OFDM system and transmitted is to be demodulated by a receiver, it is necessary to pick up synchronization information from the OFDM signal which, before attaining synchronization, looks like noise and then demodulate the signal. 
     As measures for taking synchronization between a receiver and transmitter, an example is described in JP-A-7-030513, in which no signal sections (null sections) are previously inserted into the OFDM signal periodically in the transmitter so that the null sections are used as a reference to take synchronization. 
     Further, another example is described in “A Study on Field Pickup Unit Using OFDM Modulation Scheme” by Shigeki et al, Institute of Television Engineers of Japan, Technical Report vol., 19, No. 38, pp. 7-12, August 1995, in which a group of synchronization symbols including a null section, a sweep signal (which varies from a lower limit frequency to an upper limit frequency of a transmission band during one symbol) and the like is inserted in the beginning of a frame which is a unit for data transmission processing and synchronization is taken using the synchronization symbol group. 
     The configuration of a demodulator used in the above example is shown in FIG.  2 . 
     In this demodulator, an RF signal modulated by the OFDM system is converted into an IF frequency by a receiving unit RF  90  and then converted into a baseband by a receiving unit IF  91 . The converted signal is digitized by an A/D converter  62  and is supplied to an OFDM demodulator  93 . In synchronization detection, an output signal of the receiving unit RF  90  is subjected to a square-law detection and synchronization detection in a synchronization detector  92  and an output signal of the synchronization detector  92  is supplied to the OFDM demodulator  93 . 
     JP-A-7-321762 discloses a technique in which sample values of the same synchronization symbol waveform (sweep signal) as that in a transmitter are stored in a receiver and correlation values of the sample values of the synchronization symbol waveform and sample values of a received signal are calculated in a correlation calculator, so that a clock frequency in the receiver is controlled to be coincident with a clock frequency of the received signal. 
     SUMMARY OF THE INVENTION 
     First, the prior art requires a high-frequency analog circuit for effecting the square-law detection and the synchronization detection with respect to the output signal of the receiving unit RF  90  of FIG. 2 which is an analog high-frequency signal and it is not easy to realize a stable synchronization detection circuit providing for mixing of noise. Second, when noise is mixed in the null section or multipath and fading occur in the section in which data is transmitted resulting in a reduced reception level even if the synchronization detection is digitally processed, there occurs the problem that the data transmission section is mistaken as the no signal section and finally the synchronization detection cannot be detected stably. 
     Waveforms of signals at portions of FIG. 2 are now described with reference to FIGS. 3A and 3B. 
     The output of the receiving unit RF  90  of FIG. 2 has a certain amplitude during transmission of data as shown by an OFDM received signal  71  of FIG.  3 A and the null sections of N 1 , N 2  and N 3  are provided periodically. 
     Accordingly, the OFDM demodulator  93  detects the null sections at points N 1  and N 2  of the OFDM received signal  71  of FIG. 3A to produce synchronization detection signals  72  (S 1  and S 2 ) as shown in FIG.  3 B. However, when a level is reduced due to influence of the multipath and the fading as shown by a section F 1  of the OFDM received signal  71 , a signal S 3  is produced in error. 
     Further, as shown by the null section N 3  of the OFDM received signal  71 , when noises are mixed in this null section, the null section cannot be sometimes detected as shown by the synchronization detection signal  72 . 
     Accordingly, it is an object of the present invention to provide an orthogonal frequency division multiplex data transmission apparatus capable of effecting detection of synchronization stably. 
     According to an aspect of the present invention, in order to achieve the above object, a transmitter of the data transmission apparatus using the orthogonal frequency division multiplex modulation system previously inserts a group of predetermined synchronization symbols into a transmission signal every fixed period in order to synchronize a receiver with the transmitter and the receiver demodulates a received transmission signal to a baseband OFDM signal. An absolute value of an A/D converted digital signal is taken and an obtained absolute signal is bandwidth-limited to a predetermined band width. Then, it is decided whether an amplitude of the signal is larger than or smaller than a predetermined value. A no signal period (hereinafter referred to as a null section) in the synchronization symbol group is detected on the basis of the decision result and a start point of another synchronization symbol subsequent to the null section is further detected. The time when both of the presence of the null section and the start point of the different synchronization symbol are detected is regarded as a synchronization timing and the operation timing of a demodulator for the receiver is set to the synchronization timing. 
     Further, the comparison and decision processing as to whether the amplitude of the signal converted into the absolute value and bandwidth-limited to the predetermined band width is larger than or smaller than the predetermined value is performed by using a comparator for making comparison as to whether the amplitude of the signal is larger than or smaller than the predetermined value and a counter for increasing or decreasing a count thereof in accordance with the comparison result and in the detection and decision processing of the null section, the point of time when the count is larger than or smaller than the predetermined value is detected as the start point of the null section. 
     Moreover, the detection of the start point of the another synchronization symbol subsequent to the null section is performed by using a majority decision type edge detector for deciding whether the start point is reached or not from signal states of N samples (N is an integer larger than or equal to 2) obtained from the comparison decision result. 
     In addition, the majority decision type edge detection means includes means for counting the number of times a signal of the another synchronization symbol which is produced subsequent to the null section exceeds a predetermined threshold value during a period of N samples and detecting a position of the start point of the another synchronization symbol on the basis of a count thereof and means for judging whether an arrangement of signal values of the another synchronization symbol during the period of N samples is a predetermined arrangement or not to detect the start point of the different synchronization symbol. 
     As has been described above, in the present invention, in order to detect the null section, the level of the digitized baseband OFDM signal is used to decide the null section to thereby digitize the detection processing of the null section, so that the operation is stabilized. Further, the level of the sweep signal subsequent to the null section is examined, so that the detection is hard to be influenced by noise even if noise is instantaneously mixed into the null section. 
     Further, even when multipath or fading occurs to thereby reduce the level of the received signal, the null section is confirmed by the above method and accordingly the probability of mistaking the null section can be reduced. 
     Furthermore, in the present invention, in order to synchronize the receiver with the transmitter, since the detection of the start point of the sweep signal after the null section of the received signal is not decided by only one sample and uses majority decision type edge detection in which decision is made on the basis of signal states of N samples, the start point of the sweep signal can be detected with higher accuracy even if noise is mixed into the null section of the received signal and the sweep signal subsequent to the null section. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a block diagram schematically illustrating a demodulating unit for receiver of a data transmission apparatus according to an embodiment of the present invention; 
     FIG. 2 is a block diagram schematically illustrating a demodulating unit for receiver of a conventional data transmission apparatus; 
     FIGS. 3A and 3B show waveforms of signals at portions of the demodulating unit for receiver of the conventional data transmission apparatus; 
     FIGS. 4A and 4B show waveforms illustrating timing of a synchronization symbol portion; 
     FIGS. 5A and 5B illustrate calculated results of correlation of a sweep signal; 
     FIG. 6 is a block diagram schematically illustrating an embodiment of a demodulator control unit for receiver of a data transmission apparatus according to the present invention; 
     FIG. 7 a block diagram schematically illustrating a majority decision type edge detector of the present invention; 
     FIG. 8 shows waveforms of signals at the majority decision type edge detector of the present invention; 
     FIG. 9 is a block diagram schematically illustrating another majority decision type edge detector of the present invention; 
     FIG. 10 is a block diagram schematically illustrating a transmission apparatus of an OFDM system to which the present invention is applied; 
     FIGS. 11A to  11 G show waveforms of signals at portions of a modulating unit for receiver of a data transmission apparatus according to an embodiment of the present invention; 
     FIG. 12 is a block diagram illustrating an amplitude-of-received signal decision unit; 
     FIG. 13 shows a signal level at a portion of the amplitude-of-received signal decision unit shown in FIG. 12; 
     FIG. 14 is a block diagram illustrating a null section detector; 
     FIG. 15 is a block diagram schematically illustrating a transmission unit of a data transmission apparatus of an embodiment according to the present invention; 
     FIG. 16 shows a structure of transmission data; 
     FIG. 17 shows definite examples of first and second synchronization symbols inserted into a transmission signal; 
     FIG. 18 is a diagram useful to explain operation that correlation of a reference signal contained in a receiver and a received signal is calculated to reproduce synchronization exactly; 
     FIG. 19 is a block diagram schematically illustrating a timing setting unit; 
     FIG. 20 is a block diagram illustrating a circuit arrangement of a sweep signal detector according to an embodiment of the invention; 
     FIG. 21 is a waveform diagram useful for explaining the operation of the circuit of FIG. 20; and 
     FIG. 22 is a block diagram illustrating a part of an OFDM receiver according to an embodiment of the invention, focusing on detection of the sweep signal from the received signal and synchronization of the receiver with the transmitter. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Referring first to FIG. 10, a transmitter  200  and a receiver  210  of an OFDM system to which the present invention is applied are described. In the transmitter  200 , series-arranged transmission data are supplied to a serial-parallel converter  201  to be converted into parallel-arranged transmission data, which are supplied to a reverse discrete Fourier transform circuit  202  to be reverse Fourier transformed. A synchronization signal and the like are added to the reverse Fourier transformed signal in a frequency conversion unit  203 , so that the signal is converted into a signal in a frequency band for transmission and is outputted from the transmitter  200 . 
     In the receiver  210 , the data of the transmission frequency transmitted from the transmitter  200  is converted into a baseband OFDM signal by a frequency conversion unit  211  to be subjected to processing such as synchronization detection. An output signal of the frequency conversion unit  211  is supplied to an OFDM demodulator  93  and is subjected to the discrete Fourier transform in a discrete Fourier transform circuit  212  in the OFDM demodulator  93  to produce received data. 
     Referring now to FIGS. 15,  16  and  17 , a configuration of the transmitter of a data transmission apparatus according to the present invention is described. 
     FIG. 15 schematically illustrates an internal configuration of the transmitter  200  of FIG. 10 in detail. 
     In the transmitter  200 , the series-arranged transmission data are supplied to the serial-parallel converter  201  to be converted into the parallel-arranged transmission data, which are supplied to the reverse discrete Fourier transform circuit  202  to be reverse Fourier transformed. 
     A data transmission symbol and synchronization symbol change-over unit  153  selects the transmission signal obtained by the reverse Fourier transform and a synchronization symbol produced by a synchronization symbol waveform memory  154 . FIG. 16 shows a format of the transmission signal selected by the change-over unit  153 . A frame which is a unit for data transmission includes a first synchronization symbol  161 , a second synchronization symbol  162  and a data transmission symbol  163 . A particular example of signals of the first and second synchronization symbols  161  and  162  is shown in FIG.  17 . The first synchronization symbol  161  is a null signal. The second synchronization symbol  162  is a sweep signal varying from a predetermined maximum frequency to minimum frequency. The signal produced from the change-over unit  153  is converted into an analog signal by a D/A converter  155  and is converted into a predetermined frequency by a frequency conversion unit  157  to produce a transmission signal. A local oscillator  158  and a clock generator  156  produce clocks for operating each block of the transmitter. 
     The demodulating unit for receiver according to an embodiment of the data transmission apparatus of the present invention is now described with reference to FIGS. 1 and 11. The signal transmitted from the OFDM transmitter is converted into an IF frequency by an RF/IF demodulation unit  61  to be then demodulated so that the baseband OFDM signal  71  is produced. 
     The baseband OFDM signal  71  is converted into a digital signal by the A/D converter  62 . An output signal of the A/D converter  62  is supplied to a controller  10  for demodulator. The signal supplied to the controller  10  is converted into an absolute value signal by an absolute value circuit  9 . A waveform of the absolute value signal S 10  is an OFDM signal  73  of FIG.  11 B. 
     The signal S 10  is supplied to a low pass filter  11  in order to reduce noise contained in the signal and an output signal of the low pass filter  11  is compared with a decision level set by an amplitude-of-received signal decision level setting unit  13  by an amplitude-of-received signal decision unit  12  (for example, general-purpose logic IC 74LS85) to produce a signal S 12 . 
     The amplitude-of-received signal decision unit  12  is now described with reference to FIGS. 12 and 13. 
     In FIG. 12, the output signal of the low pass filter  11  is supplied to an input terminal  110  of FIG.  12 . Further, in order to decide a magnitude of the input signal, the threshold value set by the amplitude-of-received signal decision level setting unit  13  is supplied to an input terminal  111 . The comparator  112  compares the signal supplied to the input terminal  110  with the signal supplied to the input terminal  111  and produces an “H” level signal to a decided result output  113  when the signal of the input terminal  111  is larger than the signal of the input terminal  110 . Further, the comparator  112  compares the signal of the input terminal  110  with the signal of the input terminal  111  and produces an “L” level signal to the decided result output  113  when the signal of the input terminal  111  is smaller. 
     FIG. 13 shows a relation of the input terminals  110  and  111  and the decided result output  113 . A signal level (received signal) at the input terminal  110  is represented by the abscissa axis and a signal level at the decided result output  113  is represented by the ordinate axis. When a signal level at the input terminal  111  (threshold) is set as shown in FIG. 13, the signal level at the decided result output  113  varies from “H” level to “L” level with respect to the signal level at the input terminal  111  as shown by an output waveform  114 . 
     As has been described above, the signal S 12  is low level when the signal S 12  is smaller than the set decision level and the signal S 12  is high level when the signal S 12  is larger than the decision level. A waveform of the signal S 12  is as shown by a decided result  74  of FIG.  11 C. The signal S 12  is supplied to a null section detector  14  constituted by a random walk counter (for example, general-purpose logic IC 74LS191) a value of which is increased when an input thereto is low level and is reduced when the input is high level. 
     The value of the random walk counter in the null section detector  14  is increased and reduced as shown by a count  75  shown in FIG.  11 D. 
     Operation of the random walk counter is now described with reference to FIG.  14 . The random walk counter is composed of an up/down counter portion  141  and a counter clip portion  145 . A signal inputted to an input terminal  142  is supplied to an up/down selection terminal of the up/down counter  141 . A clock for operating the up/down counter  141  is inputted to a clock terminal  143 . Whether the signal inputted to the input terminal  142  is “H” or “L” is judged at the timing of the clock inputted to the clock terminal  143 . When the signal is “H”, the value of the up/down counter  141  is increased and when the signal is “L”, the value of the up/down counter  141  is reduced. The counter clip portion  145  is supplied with an output signal  144  of the up/down counter and the input signal inputted to the input terminal  142 . When a count of the up/down counter  141  reaches a maximum value, the counter clip portion  145  controls an enable terminal EN of the up/down counter  141  to “L” so that the maximum value is held to prevent the up/down counter  141  from continuing the count-up operation and from returning to a minimum value. Further, when a count of the up/down counter  141  reaches a minimum value, the counter clip portion  145  controls the enable terminal EN of the up/down counter  141  to “L” so that the minimum value is held to prevent the up/down counter  141  from continuing the count-down operation and from returning to the maximum value. 
     When the count of the random walk counter exceeds a previously set prescribed value, the null section detector  14  judges that the section considered to be null is detected and produces a null section detection signal as shown by a waveform  76  of FIG.  11 E. 
     Further, since the signal S 10  (the OFDM signal  73  of FIG. 11B) of FIG. 1 is compared with the decision level set previously by the amplitude-of-received signal decision level setting unit  13 , predetermined synchronization signals C 1 , C 2  and C 3  are inserted after null periods N 1 , N 2  and N 3  as shown by the OFDM received signal  71  of FIG. 11A so as to be able to judge the end of null clearly. The synchronization signals C 1 , C 2  and C 3  may be the sweep signals with which the amplitude of the OFDM signal is maximized as described above. 
     A sweep signal detector  15  shown in FIG. 1 produces a signal for notifying detection of the sweep signal or the end of the null section when the sweep signal detector  15  detects the beginning (edge) of the sweep signals C 1 , C 2  and C 3 . FIG. 11F shows a waveform  77  of the end signal of the null section. 
     An operation of the sweep signal detector  15  will now be described with reference to FIGS. 20 and 21. 
     The signal S 12  in FIG. 1 which corresponds to an input signal  115  is inputted to an AND-gate  117  and is also inputted to a flip-flop  114  where it is delayed by a predetermined time. An output of the flip-flop  114  is inverted to a resultant inverted signal  116  which in turn is inputted to the AND-gate  117  to produce an output  118 . More specifically, as shown in FIG. 21, since the inverted signal  116  which is delayed relative to the input signal  115  and inverted is inputted to the AND-gate  117 , the output signal  118  of the AND-gate  117  constitutes a pulse that detects the rise of the input signal  115 , namely, the sweep signal. 
     Only when a timing setting unit or timing controller  17  for receiver shown in FIG. 1 receives the null section detection signal  76  of the null section detector  14  and the sweep signal detection signal  77  of the timing setting unit  17  for receiver in a pair, the timing setting unit  17  judges that a correct synchronization signal is received and distributes the synchronization signal shown by  79  of FIG. 11G to each block of receiver  210 . 
     FIG. 19 is a block diagram illustrating the timing controller  17 . In FIG. 19, numeral  171  denotes a one-shot multivibrator,  172  an AND gate, and  173  a counter which receives an output signal of the AND gate  172  as a reset input. The null section detection signal  76  is applied to one input of the AND gate  172  through the one-shot multivibrator  171  and the sweep signal detection signal  77  is also applied to the other input of the AND gate  172 . With this configuration, only when both the signals  76  and  77  are detected in a pair, the counter  173  outputs the synchronization signal. 
     In order to synchronize the receiver with the transmitter, the clock generator in the receiver is controlled so that the frequency and the phase of the clock of the receiver follow the frequency and the phase of the clock of the transmitter. In this control, correlation calculation of the sweep signal received by the receiver and the sweep signal provided in the receiver is made to calculate a frequency difference and a phase difference of the clocks of the receiver and the transmitter, so that the clock generator in the receiver is controlled by the calculated differences. 
     This control will be explained in detail with reference to FIG.  22 . 
     The baseband OFDM signal from the RF/IF demodulator  61  after demodulation thereat is inputted to the A/D converter  62  whose output in turn is inputted to the OFDM demodulator  93 . On the other hand, the baseband OFDM signal outputted from the A/D converter  62  is also inputted to a sync signal detector  125  where a rough sync position is detected from the received signal and is inputted to the sweep signal correlation calculation unit  127 . The sweep signal correlation calculation unit  127  receives sweep symbol data outputted from a sync symbol memory  126  and calculates correlation between the sync symbol data of the received signal and the sweep symbol data from the sync signal memory  126 . An oscillation frequency control signal generator  128  controls a local oscillator  130  in accordance with the output of the sweep signal correlation calculation unit  127  such that when the point of a sample where a peak correlation value is obtained is leading relative to a reference phase which the receiver has, the local oscillator  130  is controlled so as to decrease the oscillation frequency, while when the point of a sample where a peak correlation value is obtained is lagging relative to the reference phase of the receiver, it is controlled to increase the oscillation frequency. Based on a clock signal generated from the local oscillator  130 , a timing generator  131  generates a signal (not shown) and delivers it to respective blocks in FIG. 22, the signal being generated every frame so as to be used as a reference for the operation timing of the demodulator. The output of the local oscillator  130  is also used as a sampling clock of the A/D converter  62 . 
     Note, in the above processing, that there occurs a delay until the detected result of synchronization is produced after the output of the A/D converter  62  has been inputted to the controller  10  for demodulator due to processing of the digital signal. Accordingly, a data delay unit  18  delays the OFDM digital signal inputted to the controller  10  for demodulator in accordance with the timing controller  17  for receiver to match the output of the timing controller  17  for receiver to the phase. Outputs of the data delay unit  18  and the timing controller  17  are fed to the OFDM demodulator  93  to demodulate the OFDM digital signal. 
     In the processing that the sweep signal subsequent to the null section is detected to take synchronization, there is a case where the position of the start point of the sweep signal is judged in error due to noise mixed in the null section and the sweep signal subsequent to the null section when the start point of the sweep signal is to be detected. An example thereof is shown in FIGS. 4A and 4B. 
     Binarized synchronization symbol waveforms  1  and  2  shown in FIGS. 4A and 4B are enlarged waveforms of a section of the synchronization symbol  78  which is the synchronization symbol portion of the decision result  74  of FIG.  11 C. 
     In this connection, when the section of the synchronization symbol  78  of FIG. 11C includes the noiseless null section  1 - 1  and the noiseless sweep signal  1 - 2  as shown by the binarized synchronization symbol waveform  1  of FIG. 4A, the start point  1 - 3  of the sweep signal can be judged clearly, while when noise is mixed in the section of the synchronization symbol  78  of FIG. 11C as shown by the null section  2 - 1  and the sweep signal  2 - 2  of the binarized synchronization symbol waveform  2  of FIG. 4B, the noise  2 - 4  mixed in the null section  2 - 1  is wrongly judged as the position of the start point of the sweep signal by the simple detection of the rising of the signal. 
     As a countermeasure thereof, it is necessary to widen the range of the correlation calculation of the sweep signal and detect the exact position of the start point  2 - 3  of the sweep signal. This correlation calculation of the sweep signal is performed by a correlation calculation unit  20  of FIG.  1 . 
     Referring now to FIG. 18, the method of synchronizing the receiver with the transmitter exactly on the basis of data (symbols  161  and  162  indicating a specific time on the time axis are inserted in addition to the null symbol as shown by the data string shown in FIG. 16) transmitted by the transmitter is described. 
     An example in which the sweep symbol subsequent to the null symbol as shown in FIG.  18 ( a ) is inserted is now described. The correlation calculation unit  20  in the receiver performs the correlation calculation of a reference signal not shown and provided in the receiver and the received signal as shown in FIG.  18 ( a ). The correlation calculation is performed by shifting sample points for starting the correlation calculation in order as shown in FIG.  18 . For example, when the start point of the correlation calculation is shifted in order from k=−i to k=j, the correlation calculation results are plotted as shown in FIG.  18 ( b ). The abscissa axis k represents a sample point and the ordinate axis R represents a correlation value. In this example, since there is a sharp peak (maximum value) at k=0, the synchronization can be reproduced exactly in the receiver upon k=0. An output signal of the correlation calculation unit  20  is supplied through an OR gate  174  to the counter  173  in the timing controller  17  to reset the counter  173  to correct the shift of the start point. 
     Returning to the description of the detection of the start point of the sweep signal, in the case of the noiseless binarized synchronization symbol waveform  1  shown in FIG. 4A, since the start point  1 - 3  of the sweep signal can be detected exactly, the correlation calculation of the sweep signal is performed by j times from K=1to K=j (j is an integer larger than or equal to 2) by using the start point  1 - 3  of the sweep signal as the start point of the correlation calculation. An example of a correlation value R thereof is shown by a correlation calculation result  4  of FIG.  5 A. 
     On the other hand, in the case of the binarized synchronization symbol waveform  2  in which noise is mixed as shown in FIG. 4B, since the position of noise  2 - 4  is wrongly judged as the position of the start point of the sweep signal when the start point  2 - 3  of the sweep signal is detected, it is necessary to perform the correlation calculation of the sweep signal by i+j times from K=−i (i is an integer larger than or equal to 2) to K=j. An example of a correlation value R thereof is shown by a correlation calculation result  5  of FIG.  5 B. 
     In conclusion, when the C/N ratio of the received signal is low and noise is mixed in the null section and the sweep signal subsequent to the null section, so that the exact start point of the sweep signal cannot be detected, it is necessary that the correlation calculation of the sweep signal of a wide section in which the number of samples larger than or equal to the number of samples between the exact start point of the sweep signal and the point judged in error as the start point of the sweep signal is added is performed to detect the synchronization point, resulting in an enlargement of the scale of the correlation calculation processing of the sweep signal. 
     In view of such, in another embodiment of the present invention, a majority decision type edge detector  19  is added to the configuration of FIG. 1 in place of the sweep signal detector  15 , so that the detection accuracy of the start point of the sweep signal subsequent to the null section is further improved. 
     A configuration of a controller for demodulator of this embodiment is illustrated in FIG.  6  and operation of the majority decision type edge detector  19  is described. The baseband OFDM signal which has been transmitted by the OFDM transmitter and demodulated to be A/D converted as described above is inputted to the amplitude-of-received signal decision unit  12 . 
     The OFDM baseband signal inputted to the amplitude decision unit  12  is compared with the decision level set in the amplitude-of-received signal decision level setting unit  13  and the result signal of the comparison is supplied to the null section detector  14  and the majority decision type edge detector  19 . The null section detector  14  detects the null section and the majority decision type edge detector  19  detects the start of the sweep signal. 
     An output signal of the majority decision type edge detector  19  is supplied to the sweep signal detector  15  to detect the starting edge of the sweep signal. 
     When both of the null section detection signal of the null section detector  14  and the sweep signal detection signal of the sweep signal detector  15  are supplied to the timing controller  17  for receiver, the timing controller  17  judges that the exact synchronization is detected and distributes the synchronization signal to each block of the receiver  210  shown in FIG.  10 . 
     An embodiment of the majority decision type edge detector  19  of the present invention is now described with reference to FIG.  7 . The binarized signal compared in the amplitude-of-received signal decision unit  12  with the decision level set by the amplitude-of-received signal decision level setting unit  13  is supplied to the majority decision type edge detector  19 . 
     This signal is supplied to a shift register  21 - 1  and is shifted therefrom to shift registers  21 - 2 , . . . ,  21 -n (n is an integer larger than or equal to  2 ) successively. Output signals from the shift registers  21 - 1  to  21 -n are supplied to an adder  22 . 
     In short, an output of the adder  22  is equal to the number of samples at the time when the signals inputted to the shift registers  21 - 1  to  21 -n are larger than the decision level set in the amplitude-of-received signal decision level setting unit  13 . 
     The output of the adder  22  is compared with a set value of a sweep signal decision level setting unit  23  by the comparator  24 . For example, when it is assumed that there are 20 shift registers  21  and that the set value of the sweep signal decision level setting unit  23  is larger than or equal to 15 samples, when, of samples values of 20 samples, sample values of 15 samples or more, are larger than the decision level set in the amplitude-of-received signal decision level setting unit  13 , it is judged that there is a start point of the sweep signal. 
     As has been described above, since the binarized signal compared in the amplitude-of-received signal decision unit  12  with the decision level set in the amplitude-of-received signal decision level setting unit  13  is edge-detected by the majority decision type edge detector  19 , influence of noise mixed in the null section of the received signal and the sweep signal subsequent to the null section is reduced as compared with the configuration of FIG.  1 . 
     Further, for example, when the sweep signal decision level setting unit  23  of FIG. 7 is set as “when, of sample values of 20 samples, the levels of 15 samples or more of the received signals are larger than the decision level set in the amplitude-of-received signal decision level setting unit  13 ”, the start point of the sweep signal is actually detected within the range from 15 samples to 20 samples of sample values from among the 20 samples and accordingly the detection timing of the start point of the sweep signal is shifted within the range of maximum 5 samples depending upon the quantity of mixed noise. 
     Since the shift of 5 samples is a shift of the detection position of the start point of the sweep signal, the correlation calculation of the sweep signal shown in FIG. 5 is required to be performed within a wide range containing the shift of 5 clocks as described above. 
     This point is now described with reference to FIG. 8 showing a concrete operation timing within the majority decision type edge detector  19 . 
     When 20 shift registers  21  are provided in the FIG. 7 arrangement and the noiseless sweep signal as shown by an input waveform  30  of FIG. 8 is supplied to the shift registers  21 , the comparator  24  judges whether the output value of the adder  22  is larger than or equal to  15  set in the sweep signal decision level setting unit  23  or not and produces an output waveform  31 . A delay period  41  at this time is 15 samples. Further, the output of the shift register  21 -n is an output waveform  33 . 
     On the other hand, when a signal mixed with noise of four samples as shown by an input waveform  34  of FIG. 8 is supplied to the shift registers  21 , the comparator  24  produces an output waveform  35  and a delay period  44  is 19 samples. 
     As has been described above, the output timing of the comparator  24  is varied in accordance with the quantity of noise and accordingly when the output of the adder  22  is judged as the start point of the sweep signal on the basis of this condition as it is, four-sample error occurs as shown by detection error  43 . 
     Accordingly, an edge decision unit  25  of FIG. 7 examines the output of the comparator  24  and the state of the shift register  21 -n (four samples in this example) and makes decision at the timing when the levels thereof are L, L, H and H, for example. 
     More particularly, the states of the samples marked with a circle as shown by the output waveform  33  or  36  of FIG. 8 are judged. 
     Consequently, even if noise is mixed in the null section of the received signal and the sweep signal subsequent to the null section, the start point of the sweep signal can be detected exactly and accordingly the range of correlation calculation can be narrowed. 
     Another embodiment of the majority decision type edge detector of the present invention is now described with reference to FIG.  9 . 
     The signal compared in the amplitude-of-received signal decision unit  12  of FIG. 1 with the decision level set by the amplitude-of-received signal decision level setting unit  13  and binarized is supplied to the majority decision type edge detector  19 . 
     This signal is supplied to the shift register  21 - 1  and is shifted therefrom to the shift registers  21 - 2 , . . . ,  21 -n successively. An adder  22 - 1  adds inputted binarized decision values and the addition result thereof is stored in a register  26 - 1 . 
     On the other hand, the outputs of the shift register  21 -n are added in an adder  22 - 2  and the addition result thereof is stored in a register  26 - 2 . 
     An adder  22 - 3  subtracts an output of the register  26 - 2  from an output of the register  26 - 1  so that the number of those samples produced between from the shift registers  21 - 1  to  21 -n that are larger than or equal to the decision level set by the amplitude-of-received signal decision level setting unit  23  is obtained. 
     The comparator  24  compares this value with the number of samples set by the sweep signal decision level setting unit  23  and decides that there is a sweep signal when the value is larger than or equal to the number of samples set by the sweep signal decision level setting unit  23 . However, since the output of the comparator  24  is shifted in the detection timing in accordance with the quantity of mixed noise as shown by the output waveforms  33  and  36  of FIG. 8 in the same manner as the comparator  24  of the majority decision type edge detector  19  of FIG. 7, the output of the comparator  24  and the state of the shift register  21 -n (four samples in this example) are examined and decision is made at the timing that the levels thereof are L, L, H and H, for example in the same manner as the edge detector  25  of FIG.  7 . 
     In the case of the majority decision type edge detector  19 , since the registers  26 - 1  and  26 - 2  have addition loops, initialization thereof is required. The initialization is made by a reset signal  27 . Further, it is necessary that the number of bits of the registers  26 - 1  and  26 - 2  can express the number larger than or equal to the number of total stages of the shift registers  21 - 1  to  21 -n. 
     In the above example, while the sweep signal subsequent to the null section is used as the synchronization symbol group, any signals having a fixed amplitude, such as a signal containing only one carrier signal having a fixed level or the like may be used as the signal subsequent to the null section. 
     In the present invention, even when the detection of the null section fails due to noise mixed in the null section or when the multipath or fading occurs to reduce the level of the received signal, the probability of mistaking the start point of the sweep signal as a null section is reduced and since the detection processing of the null section is digitized, there can be provided a stable OFDM transmission apparatus capable of exactly detecting the start point of the sweep signal.