Abstract:
A method and an apparatus for monitoring the wall signal input to the wall filter of a spectral Doppler processor to check for probe-motion-induced clutter. This clutter is typically of higher frequency and amplitude than that due to normal vessel wall motion. Some additional threshold logic is used to check for energy within a frequency band greater than the normal wall signal frequencies. If significant energy above some “rattle” threshold is detected for a predefined time interval, the Doppler audio is automatically muted. This can be effected at one or more points within the normal Doppler audio signal path in a conventional scanner. If the rattling clutter is no longer detected, the Doppler audio is re-activated or ramped up smoothly.

Description:
RELATED PATENT APPLICATION 
     This application is a continuation-in-part application claiming priority from U.S. patent application Ser. No. 09/349,586 filed on Jul. 9, 1999, U.S. Pat. No. 6,296,612. 
    
    
     FIELD OF THE INVENTION 
     This invention relates to ultrasonic diagnostic systems which measure the velocity of blood flow using spectral Doppler techniques. In particular, the invention relates to the continuous display of such information, including maximum and mean blood flow velocities. 
     BACKGROUND OF THE INVENTION 
     Ultrasonic scanners for detecting blood flow based on the Doppler effect are well known. Such systems operate by actuating an ultrasonic transducer array to transmit ultrasonic waves into the object and receiving ultrasonic echoes backscattered from the object. For blood flow measurements, returning ultrasonic waves are compared to a frequency reference to determine the frequency shifts imparted to the returning waves by moving objects including the vessel walls and the red blood cells inside the vessel. These frequency shifts translate into velocities of motion. 
     In state-of-the-art ultrasonic scanners, the pulsed or continuous wave Doppler waveform is computed and displayed in real-time as a gray-scale spectrogram of velocity versus time with the gray-scale intensity (or color) modulated by the spectral power. The data for each spectral line comprises a multiplicity of frequency data bins for different frequency intervals, the spectral power data in each bin for a respective spectral line being displayed in a respective pixel of a respective column of pixels on the display monitor. Each spectral line represents an instantaneous measurement of blood flow. 
     In the conventional spectral Doppler mode, an ultrasound transducer array is activated to transmit by a transmit ultrasound burst which is fired repeatedly at a pulse repetition frequency (PRF). The PRF is typically in the kilohertz range. The return radiofrequency (RF) signals are detected by the transducer elements and then formed into a receive beam by a beamformer. For a digital system, the summed RF signal from each firing is demodulated by a demodulator into its in-phase and quadrature (I/Q) components. The I/Q components are integrated (summed) over a specific time interval and then sampled. The summing interval and transmit burst length together define the length of the sample volume as specified by the user. This so-called “sum and dump” operation effectively yields the Doppler signal backscattered from the sample volume. The Doppler signal is passed through a wall filter, which is a high pass filter that rejects any clutter in the signal corresponding to stationary or very slow-moving tissue, including a portion of the vessel wall(s) that might be lying within the sample volume. The filtered output is then fed into a spectrum analyzer, which typically takes the complex Fast Fourier Transform (FFT) over a moving time window of 64 to 256 samples. The data samples within an FFT analysis time window will be referred to hereinafter as an FFT packet. The FFT output contains all the information needed to create the video spectral display as well as the audio output (typical diagnostic Doppler ultrasound frequencies are in the audible range). 
     For video display, the power spectrum is computed by taking the power, or absolute value squared, of the FFT output. The power spectrum is compressed and then displayed via a gray-scale mapping on the monitor as a single spectral line at a particular time point in the Doppler velocity (frequency) versus time spectrogram. The positive frequency [0:PRF/2] spectrum represents flow velocities towards the transducer, whereas the negative frequency [−PRF/2:0] spectrum represents flow away from the transducer. An automatic Doppler maximum/mean waveform tracing is usually performed after the FFT power spectrum has been compressed. The computed maximum/mean velocity traces are usually presented as overlay information on the spectrogram display. 
     For the audio Doppler output, the positive and negative frequency portions, or sidebands, of the FFT output are split into two separate channels representing the forward and reverse flow spectra respectively. For each channel, the sideband is reflected about the zero frequency axis to obtain a symmetric spectrum, which generates, after an inverse FFT (IFFT) operation, a real-valued flow signal in the time domain. Both the forward and reverse flow signals are converted into analog waveforms, which are fed to corresponding audio speakers. 
     During a spectral Doppler exam, the sonographer often needs to move the probe over an anatomical region surrounding some vascular system. Probe motion effects may also result simply from large tissue movements due to breathing or other body motion. Whenever the Doppler sample volume is being jerked around over body tissue, low-frequency clutter is generated which can be significantly stronger than vessel wall signals. Such probe-motion-induced clutter often exceeds the wall filter cutoff frequency, and will show up as blooming white (very strong) echoes right above the wall filter region in the spectral display. This can be considered as the audio counterpart of the “flash artifact” in color flow imaging. The annoying effect lies not so much in the video display, but in the Doppler audio: such clutter usually generates a loud rattle that may scare the patient and/or increase his/her anxiety level. 
     Some form of automatic gain control is usually available in conventional Doppler scanners. Automatic gain control is used to prevent large signals from saturating various points in the signal chain including the video display and/or audio speaker output. Automatic gain control generally consists of detecting the signal amplitude level and adjusting the gain down if the signal approaches a maximum allowable level. Such gain control does not specifically attempt to detect the presence of and to completely mute out the loud probe-motion-induced clutter noise. 
     There is a need for a Doppler processor that can monitor the I/Q data for the presence of probemotion-induced clutter before the I/Q data is wall filtered. 
     SUMMARY OF THE INVENTION 
     The present invention is a method and an apparatus for monitoring the wall signal input to the wall filter of a spectral Doppler processor to check for probe-motion-induced clutter. This clutter is typically of higher frequency and amplitude than that due to normal vessel wall motion. In accordance with the preferred embodiment of the invention, some additional threshold logic is used to check for energy within a frequency band greater than the normal wall signal frequencies. If significant energy above some “rattle” threshold is detected for a predefined time interval, the Doppler audio is automatically muted. This can be effected at one or more points within the normal Doppler audio signal path in a conventional scanner. If the rattling clutter is no longer detected, the Doppler audio is re-activated or ramped up smoothly. 
     In accordance with the preferred embodiment, a system noise model is used to predict the mean system noise power in a bandpass filter output. The mean system noise power predicted by the system noise model provides a noise threshold to gage how much probe-motion-induced clutter power is present in the current FFT packet. If no significant probe-motion-induced clutter is present, then the audio processing will not be turned off. If significant probe-motion-induced clutter power is present in the FFT packet, the audio processing is turned off. 
     It should be clear to those skilled in the art that the method of the invention can be implemented in hardware (e.g., a digital signal processing chip) and/or software. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a block diagram of the basic signal processing chain in a conventional spectral Doppler imaging system. 
     FIG. 2 is a block diagram showing an automatic audio muting mechanism in accordance with the preferred embodiment of the invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     A typical digital real-time ultrasonic imaging system having a spectral Doppler imaging mode is generally depicted in FIG.  1 . An ultrasound transducer array  2  is activated to transmit by a transmit ultrasound burst which is fired repeatedly at a pulse repetition frequency (PRF). The PRF is typically in the kilohertz range. The return RF signals are detected by the transducer elements and then formed into a receive beam by a beamformer  4 . For a digital system, the summed RF signal from each firing is demodulated by demodulator  6  into its in-phase and quadrature (I/Q) components. The I/Q components are integrated (summed) over a specific time interval and then sampled by a “sum and dump” block  8 . The summing interval and transmit burst length together define the length of the sample volume as specified by the user. The “sum and dump” operation effectively yields the Doppler signal backscattered from the sample volume. The Doppler signal is passed through a wall filter  10  which rejects any clutter in the signal corresponding to stationary or very slow-moving tissue. The filtered output is then fed into a spectrum analyzer comprising a Fast Fourier Transform (FFT) block  12  and a power computation block  14 . The FFT block  14  performs Fast Fourier transformation over a moving time window of 64 to 256 samples. Each FFT power spectrum output by block  14  is compressed (block  16 ) and sent to a known display system  18  comprising a timeline display memory, a video processor and a display monitor. The video processor maps the compressed FFT power spectral data to a grayscale for display on the monitor as a single spectral line at a particular time point in the Doppler velocity (frequency) versus time spectrogram. 
     For the audio Doppler output, the positive and negative frequency portions, or sidebands, of the output of the FFT block  12  are split by a sideband splitter  20  into two separate channels representing the forward and reverse (designated “FWD.” and “REV.” in FIG. 1) flow spectra respectively. For each channel, the sideband is reflected about the zero frequency axis to obtain a symmetric spectrum, which generates, after an inverse FFT (IFFT) operation (block  22 ), a real-valued flow signal in the time domain. Both the forward and reverse flow signals are converted into analog waveforms by respective digital-to-analog converters (DACs)  24 . The analog waveforms are fed to corresponding audio speakers  26 . 
     In accordance with the preferred embodiment of the invention, generally depicted in FIG. 2, an automatic audio muting mechanism receives the same signal which is input to the wall filter  10 . In this particular embodiment, the audio muting mechanism is placed in parallel with the wall filter  10  and FFT block  12 . The wall filter and FFT blocks are the same as those of a conventional scanner (shown in FIG.  1 ). The primary additional processing steps in accordance with the preferred method of practicing the invention are detailed below. It should be apparent that all the functional blocks shown in FIG. 2 can be implemented in software and/or hardware. 
     In accordance with the preferred embodiment, the I/Q data samples from the “sum and dump” block  8  are bandpass filter  28  is designed to pass a band of frequencies that are typical of probe motion, and to reject wall signals produced by slower-moving vessels and/or signals produced by very fast-moving blood flow. The instantaneous power of the filtered signal is then computed in block  30 . The instantaneous power of the filtered signal is given by I 2 +Q 2 , which is a measure of the strength of the reflected signal. Preferably, the total power of the bandpass filter output, i.e., the sum of (I n   2 +Q n   2 ) over the FFT (or a fraction of the FFT) packet size M, is computed. A threshold logic block  32  checks if the signal power has exceeded a predefined “rattle” threshold. This threshold test may also be based on a moving average of the power over a predefined time interval, such as 20 msec. If the instantaneous or integrated power exceeds the threshold, the threshold logic block  32  will issue a flag to turn off the audio processing (i.e., mute the audio output). In the block diagram of FIG. 2, such a flag is equivalent to sending a “0” value from the threshold logic block  32  to a multiplier  36  situated in the audio path, i.e., situated in the line connecting the output of the FFT block  12  to the input to the sideband splitter  20  (see FIG.  1 ). If the audio is in a muted state, and the instantaneous or integrated power falls below the tolerance level, the threshold logic block  32  issues a “1” to the multiplier  36  to turn on the audio processing again, or it may issue a slow ramp-up function, e.g., a signal which increases linearly from “0” to “1”, to gradually turn the audio back on. 
     As shown in FIG. 2, the threshold level may be derived from a system noise model  34  which can predict the system noise level given the current system setup and front-panel settings (e.g., the Doppler gain setting). This is a known art, especially for digital systems in which the main noise sources lie in the front-end analog electronics. The noise model may be implemented as a processor programmed to calculate a set of equations that predict the system noise, or in the form of a look-up table with pre-calculated values. The “rattle” threshold employed by the threshold logic block  32  is set at some predetermined level (i.e., threshold) above the noise level. That is, if the signal power exceeds this threshold, then it is considered to be noise caused by probe motion and the corresponding audio output will be quite annoying if it is not muted. 
     In accordance with the preferred embodiment, the system noise model  34  is used to predict the mean power of the system noise within the passband of the bandpass filter  28 . In the most preferred embodiment, the model assumes an all-digital scanner whose system noise originates primarily from the pre-amplifier in each receive channel in the beamformer. The pre-amp Johnson noise is often specified as arms voltage per Hz ½  (e.g., 0 nV/Hz ½ ) at room temperature. Thus, knowing the equivalent noise bandwidths of all the filters in the Doppler signal path (from the demodulator to the “sum and dump” filter) should enable an absolute arms noise level to be computed as a function of system gain. Any quantization noise due to analog-to-digital conversion in the receiver can also be added in an appropriate manner. Further, knowing the sample volume position and aperture strategy in the spectral Doppler mode, it should be straightforward to compute the total system noise by summing over all active receive channels (including array apodization effects) for a given sample volume position. The mean noise power at the bandpass filter output can be computed based on the bandwidth of the bandpass filter  28 . It should be apparent to those skilled in the art that similar noise models can be developed for scanners whose Doppler signal paths differ from the basic structure of FIG.  1 . Also, while a system noise model is clearly most efficient from an implementation standpoint, a LUT with multiple inputs can be used to perform the same function. Such a LUT can be established either by noise calibration measurements or by simulating the system noise model. In the first alternative, the system is pre-calibrated by trying different combinations of gain settings, recording the resulting noise values and storing those gain settings and corresponding noise values in a LUT. In the second alternative, the noise model values are pre-computed and stored in a LUT. 
     While the invention has been described with reference to preferred embodiments, it will be understood by those skilled in the art that various changes may be made and equivalents may be substituted for elements thereof without departing from the scope of the invention. In addition, many modifications may be made to adapt a particular situation to the teachings of the invention without departing from the essential scope thereof. Therefore it is intended that the invention not be limited to the particular embodiment disclosed as the best mode contemplated for carrying out this invention, but that the invention will include all embodiments falling within the scope of the appended claims. 
     As used in the claims, the term “digital signal processor” includes digital signal processing hardware and/or software.