Abstract:
Where two or more multi-valued digital data symbols are modulated so that they overlap after passing through a channel, forming a combined signal, a receiver receives the combined signal and forms detection statistics to attempt to recover the symbols. Where forming detection statistics does not completely separate the symbols, each statistic comprises a different mix of the symbols. A receiver determines the symbols which, when mixed in the same way, reproduce or explain the statistics most closely. For example, the receiver hypothesizes all but one of the symbols and subtracts the effect of the hypothesized symbols from the mixed statistics. The remainders are combined and quantized to the nearest value of the remaining symbol. For each hypothesis, the remaining symbol is determined. A metric is then computed for each symbol hypothesis including the so-determined remaining symbol, and the symbol set producing the best metric is chosen as the decoded symbols.

Description:
RELATED APPLICATIONS 
       [0001]    This application claims priority under 35 U.S.C. § 120 as a continuation-in-part of U.S. patent application Ser. No. 12/035,846, entitled “A Method and Apparatus for Block-Based Signal Demodulation,” as filed on 22 Feb. 2008, the entire contents of which are incorporated herein by reference. 
     
    
     TECHNICAL FIELD 
       [0002]    The invention generally relates to communication signal processing, and particularly relates to simplifying the detection of digital data symbols transmitted, for example, using multiple, code division multiple access (CDMA) codes as used in the Third Generation (3G) of mobile wireless communications systems. 
       BACKGROUND 
       [0003]    Mobile wireless communications systems have evolved from principally telephone voice services to provide higher data rates for mobile Internet browsing, MP3 music or TV and video. Transmission techniques are varied and include mixtures of FDMA, TDMA and CDMA. In the latter, a data stream is modulated upon a high chip-rate CDMA code that is preferably orthogonal to the codes used by other data streams, allowing the receiver to separate out its desired data stream from the others by correlating with a particular code. If the codes are truly orthogonal, correlation, also known as despreading, should separate out the desired data stream with complete cancellation of the other, unwanted data streams. However, orthogonal codes do not remain perfectly orthogonal when relatively delayed. Relatively delayed copies of the signal are received due to being reflected from large objects such as buildings and mountains, giving rise to multipath propagation. 
         [0004]    To increase data rate to a given receiver, the receiver may be assigned to receive multiple data streams using different orthogonal codes. In one 3G system known as HSPA, the highest uplink data rate is provided by assigning three-quarters of the entire code space to one receiver. The standard for this system specifies assigning one code of spreading factor 2 to the receiver, which represents half the code space, and one code of spreading factor four, which represents half of the remaining code space. Using these codes, in a period of four CDMA chips, two spreading factor=2 symbols are transmitted overlapping one spreading factor=4 symbol, giving three data symbols per 4 code chips. Moreover, the symbols can each be 16QAM symbols of four bits each. A conventional “brute-force” detector for three such 16QAM symbols would hypothesize all 16 3 =4096 possibilities and determine the symbol triple that best explained the signal that was received through the multipath channel. The multipath channel characteristics are separately determined using the technique of channel estimation. However, testing 4096 hypotheses is onerous on signal processing circuits and thus an improved method and apparatus for determining the most likely received symbol triples is required. 
         [0005]    A similar joint detection problem occurs in non-spread systems, such as GSM/EDGE and Long Term Evolution (LTE). In a cellular system, multiple users can be assigned the same channel resource (frequency subcarrier or time slot). Receiver performance can be improved by jointly detecting such users. Brute-force detection would require hypothesizing all combinations of the symbols from the multiple users. In the LTE uplink, self-interference can also be a problem, which can be alleviated by jointly detecting symbols in a block equalizer, such as a block decision feedback equalizer. 
       SUMMARY 
       [0006]    Two or more multi-valued digital data symbols are modulated onto the same channel resource. A receiver receives the combined signal and forms detection statistics to attempt to recover the symbols. Due either to the symbol waveforms being non-orthogonal, or due to multipath distortion of orthogonal waveforms, the detection statistics do not completely separate the symbols with the result that each statistic comprises a different mix of the symbols. A receiver configured according to the teachings presented herein determines the symbols which, when mixed in the same way, reproduce the statistics most closely. In one or more embodiments, the inventive method comprises hypothesizing all but one of the symbols and subtracting the effect of the hypothesized symbols from the mixed statistics. The remainders are then combined and quantized to the nearest value of the remaining symbol. Thus for each hypothesis of all but one of the symbols, the remaining symbol is determined. A metric is then computed for each symbol hypothesis including the so-determined remaining symbol and the symbol set that produces the best metric is chosen as the decoded symbols. 
         [0007]    However, the present invention is not limited to the above features and advantages. Indeed, those skilled in the art will recognize additional features and advantages upon reading the following detailed description, and upon viewing the accompanying drawings. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0008]      FIG. 1  is a diagram of known orthogonal coding, as may be used to encode multiple symbols into a symbol block. 
           [0009]      FIG. 2  is a diagram of a known 16QAM constellation. 
           [0010]      FIG. 3  is a diagram of known symbol coding, where three symbols are formed using length=2 and length=4 coding for combining in a symbol block. 
           [0011]      FIG. 4  is a diagram of an example symbol block structure as might arise using the coding of  FIG. 3 , such as is contemplated for processing via efficient multi-symbol detection as taught herein. 
           [0012]      FIG. 5  is a diagram of another example symbol block structure, such as is contemplated for processing via efficient multi-symbol detection as taught herein. 
           [0013]      FIG. 6  is a block diagram of one embodiment of a receiver apparatus configured for efficient multi-symbol detection as taught herein. 
           [0014]      FIG. 7  is a block diagram of one embodiment of the block equalizer illustrated in  FIG. 6 . 
           [0015]      FIG. 8  is a block diagram of another embodiment of a receiver apparatus configured for efficient multi-symbol detection as taught herein. 
           [0016]      FIG. 9  is a flow diagram of one embodiment of processing logic for implementing a method of efficient multi-symbol detection as taught herein. 
       
    
    
     DETAILED DESCRIPTION 
       [0017]      FIG. 1  illustrates a set of four orthogonal codes,  1010 ,  1100 ,  0110 , and  1111 . The codes are orthogonal when as many bits agree as disagree. When these Boolean code representations are used to generate real signals, the Boolean values (1, 0) are usually mapped to arithmetic signal values (−1, 1) or the reverse.  FIG. 2  illustrates a 16QAM constellation with Grey Code mapping for representing groups of four bits by a signal point in the complex plane with real and imaginary parts each taking on one of the four values +/−0.5 or +/−1.5. An alternative expression for the transmitted signal thus could be (b 1 +b 2 /2)+j(b 3 +b 4 /2). However, the mapping of four-bit groups to signal points is preferably Grey Code mapping of bit quadruples to signal points such that adjacent patterns, which are most likely to be confused due to noise, differ in only one bit and not two. However, if the bits b 1 , b 2 , b 3 , and b 4  in the above expression can be determined, a symbol is identified, the 4-bit Grey code assigned to that symbol used as the decoded data. 
         [0018]    A CDMA signal modulated with 16QAM symbols is obtained by choosing any code from  FIG. 1  and multiplying each bit of it with the symbol value of the 4-bit group to be transmitted, as given by  FIG. 2 . Thus, in principle four 16QAM symbols could be transmitted in parallel by using all four codes of  FIG. 1  to carry respective 16QAM symbols. After adding, the composite signal is often scrambled by applying a common, complex scrambling code which has the purpose of distinguishing transmissions from different transmitters using the same frequency. A receiver for such a signal could comprise receiving four signal samples r 1 , r 2 , r 3 , and r 4  corresponding to the four code chips (after removing the above-mentioned, common, complex scrambling code) and forming R 1 =r 1 +r 2 +r 3 +r 4  (correlation with code  1111 ); R 2 =r 1 −r 2 +r 3 −r 4  (correlation with code  1010 ); R 3 =−r 1 +r 2 +r 3 −r 4  (correlation with code  0110 ); and R 4 =r 1 +r 2 −r 3 −r 4  (correlation with code  1100 ). Such correlations or detection statistics are often most simply computed by performing a Fast Walsh Transform (FWT) on the received values, after first having removed any common complex scrambling code. 
         [0019]    The R&#39;s thus derived from the received signal samples r 1 , r 2 , r 3 , and r 4  may be regarded as modified received signal samples to be used for further processing. Indeed, a formulation of the present invention based on this example correlation receiver will first be derived, but other types of chip-level preprocessing receivers such as a RAKE receiver, matched filtering, decision feedback equalization (DFE) or other equalizing receiver and noise whitening receivers could be employed, all of which, according to the present invention, still give rise to equations of the same form as Eq. (1) to Eq. (4) below. 
         [0020]    In this sense, it should be understood that the teachings herein apply directly to CDMA and non-CDMA received signals. For example, efficient multi-symbol detection as taught herein may be applied to the types of direct-sequence (DS) CDMA signals used in Wideband CDMA (WCDMA) based communication systems, but also may be readily applied to received signal processing in communication systems based on the Long Term Evolution (LTE) standards, as promulgated by the Third Generation Partnership Project (3GPP). For example, efficient multi-symbol detection as taught herein advantageously applies to uplink signal processing in LTE systems. 
         [0021]    Turning back to processing details, if code orthogonality is preserved in transmission, the resulting modified received signal values R 1 , R 2 , R 3 , R 4  are simply equal to some channel constant Q times the original 16QAM signal values. More particularly, the Q-values determine how each R-value depends on each symbol value (S-value) in the block of symbol values to be detected. Thus, the Q-values may be termed dependence coefficients. Dividing by Q (or multiplying by Q* and dividing by |Q| 2 ) reproduces the symbol values except for added noise. Quantizing these “soft” symbol values to the nearest of the possible symbol values in the alphabet then gives the decoded symbol. The real and imaginary parts can be quantized separately. Quantizing the imaginary part to one of −1.5, −0.5, +0.5, and +1.5 yields the first two bits of the bit quadruple, and quantizing the real part likewise yields the other two bits. Thus with perfect orthogonality and no noise, we expect 
         [0000]        Q·S 1= R 1, 
         [0000]        Q·S 2= R 2, 
         [0000]        Q·S 3= R 3, and 
         [0000]        Q·S 4= R 4. 
         [0000]    With imperfect orthogonality however, R 1  will be corrupted by some amount of S 2 , S 3  and S 4 , and likewise for the other correlation values, with the result given by equation 1 below: 
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         [0000]    The Q-values determine how each R-value depends on each S-value, and so may be termed dependence coefficients. 
         [0022]    Attempting to solve such equations by inverting the 4×4 Q-matrix is not advisable, as it will in general result in magnifying the receive noise present on the R-values. Instead, sets of symbols (S 1 , S 2 , S 3 , S 4 ) may be hypothesized, multiplied by the Q matrix, and the results compared for a match to the values (R 1 , R 2 , R 3 , R 4 ). The symbol set producing the best match is then selected. The latter method produces a more accurate result because the symbols are constrained to their discrete values, unlike matrix inversion, which does not produce discrete values. However, hypothesizing four 16QAM symbols means computing and testing 16 4 =65,536 metrics, or, in general, M N  metrics, where N is the number of symbols and M is the size of the symbol alphabet. We now disclose how the number of metrics to be computed can be reduced to M N−1 , or 4096 tests in the above 16QAM example. 
         [0023]    First, remove every symbol except one from the left-hand-side (LHS) of the above equations to the right-hand-side (RHS), obtaining, for example: 
         [0000]        Q 11· S 1= R 1− Q 12· S 2− Q 13· S 3− Q 14· S 4= R 1′, 
         [0000]        Q 21· S 1= R 2− Q 22· S 2− Q 23· S 3− Q 24· S 4= R 2′, 
         [0000]        Q 31· S 1= R 3− Q 32· S 2− Q 33· S 3− Q 34· S 4= R 3′, and 
         [0000]        Q 41· S 1= R 4− Q 42· S 2− Q 43· S 3− Q 44· S 4= R 4′.  Eq. (2) 
         [0000]    Thus the dependence of the R-values on the values S 2 , S 3 , and S 4  is removed. Now, on the assumption that receiver noise on each R-value has the same RMS value and are uncorrelated with each other, the best value of S 1  is found by weighting each equation by the conjugate of the Q-value on the left, and adding, obtaining [|Q 11 | 2 +|Q 21 | 2 +|Q 31 | 2 +|Q 41 | 2 ]S 1 =[Q 11 *R 1 ′+Q 21 *R 2 ′+Q 31 *R 3 ′+Q 41 *R 4 ′]. Thus, 
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         [0000]    The above is the “optimum diversity combiner” solution for determining S 1 , given the four independent equations. The result is a “soft” value for S 1  (e.g., an unquantized value), which must now be quantized to the nearest discrete symbol value. The real and imaginary parts may be quantized separately, as follows: (i) determine the sign; and (ii) determine if the magnitude is greater or less than 1. This quantizing operation determines, for the 16QAM example, whether the value is nearest to −1.5, −0.5, +0.5 or +1.5, as required. 
         [0024]    The above process is carried out for each of the 4096 possible combinations of the symbol triples (S 2 , S 3 , S 4 ) and the resulting quantized value of S 1  is associated with each to obtain a candidate symbol quadruple (S 1 , S 2 , S 3 , S 4 ). Instead of testing 65,536 possibilities, it is now only necessary to test which of the 4096 candidates best explains the received signal sample values. This is done by computing for each a metric which is the sum of the squares of the magnitudes of the mismatch between the LHS and RHS of the equations when the candidate symbol set is substituted, and choosing the candidate symbol set giving the lowest metric. 
         [0025]    Much reduction of computation is possible by noting that Eq. (3) may be written 
         [0000]        S 1= R″−Q 2′ ·S 2 −Q 3 ′·S 3 −Q 4 ′·S 4  Eq. (4) 
         [0000]    where 
         [0000]      R″=[Q 11 *R 1 +Q 21 *R 2 +Q 3 *R 3 +Q 41 *R 4 ]/A, 
         [0000]    where 
         [0000]      A=[|Q 11 | 2 +|Q 21 | 2 +|Q 31 | 2 +|Q 41 | 2 ], and 
         [0000]      S Q 2 ′=[Q 11 *Q 12 +Q 21 *Q 22 +Q 31 *Q 32 +Q 41 *Q 42 ]/A, 
         [0000]      Q 3 ′=[Q 11 *Q 13 +Q 21 *Q 23 +Q 31 *Q 33 +Q 41 *Q 43 ]/A, and 
         [0000]      Q 4 ′=[Q 11 *Q 14 +Q 21 *Q 24 +Q 31 *Q 34 +Q 41 *Q 44 ]/A, 
         [0000]    all of which may be pre-computed only once. R″ is a weighted combination of the original signal samples and Q 2 ′, Q 3 ′ and Q 4 ′ may be termed modified dependence coefficients. 
         [0026]    Thus Eq. (4) describes removing from the combined signal value R″ the dependence on S 2 , S 3 , and S 4  by using the modified dependence coefficients. The value R″ may be equated to a matched filtered received value, as often arises in equalizers using an Ungerboeck metric formulation. Furthermore, there is a 4-fold rotational symmetry in the term −Q 2 ′S 2 −Q 3 ′S 3 −Q 4 ′S 4  which allows the equation to be computed for values of S 2  only in the first quadrant, and then values with S 2  in the other quadrants are determined by rotating the result through 90, 180 and 270 degrees, all of which require only exchange of real and imaginary parts and sign changes. 
         [0027]    Once the candidate symbol groups (S 1 , S 2 , S 3 , S 4 ) have been identified as above, testing to see which best fulfills Eq. (1) may be simplified as follows. The sum-square error in satisfying the four included equations is the metric to be minimized, and is equal to: 
         [0000]      |Q 11 ·S 1 +Q 12 ·S 2 +Q 13 ·S 3 +Q 14 ·S 4 −R 1 | 2 +|Q 21 ·S 1 +Q 12 ·S 2 +Q 23 ·S 3 +Q 24 ·S 4 −R 2 | 2 |Q 31 ·S 1 +Q 32 ·S 2 +Q 33 ·S 3 +Q 34 ·S 4 −R 3 | 2 +|Q 41 ·S 1 +Q 42 ·S 2 +Q 43 ·S 3 +Q 44 ·S 4 −R 4 | 2 .  Eq. (5) 
         [0000]    The sub-terms such as Q 12 ·S 2 +Q 23 ·S 3 +Q 24 ·S 4  are required to be computed for all 4096 combinations of S 2 , S 3 , and S 4 . However, such terms may be split into real and imaginary parts, by letting Q=(U+jV) and S=(X+jY) with corresponding indices, thus obtaining: 
         [0000]        Q 12 ·S 2+ Q 23 ·S 3+ Q 24 ·S 4=[( X 2 ·U 12 +X 3 ·U 23 +X 4 ·U 24)−( Y 2 −V 12+ Y 3 ·V 23+ Y 4 ·V 24)]+ j [( X 2 ·V 12 +X 3 −V 23+ X 4 ·V 24)+( Y 2 −U 12+ Y 3 ·U 23+ Y 4 ·U 24)] 
         [0028]    Letting 
         [0000]        T 1( X 2, X 3, X 4)=( X 2· U 12+ X 3· U 23+ X 4· U 24), 
         [0000]        T 2( Y 2, Y 3, Y 4)=( Y 2· V 12+ Y 3· V 23+ Y 4· V 24), 
         [0000]        T 3( X 2, X 3, X 4)=( X 2· V 12+ X 3· V 23+ X 4· V 24), and 
         [0000]        T 4( Y 2, Y 3, Y 4)=( Y 2· U 12+ Y 3· U 23+ Y 4· U 24), 
         [0000]    it may be noted that there are only 8 values of each of T 1 , T 2 , T 3  and T 4  to be computed, as the X and Y are only 2-bit values. These may be further arranged so that no multiplications are required. Thus, by pre-computing the 4×8 T-values, terms such as Q 12  S 2 +Q 23  S 3 +Q 24 . S 4  may be computed just by adding or subtracting two of the T-values to obtain each of the real and the imaginary parts. Likewise the term such as Q 31 ·S 1  is required for all 16 values of S 1 , but computing imaginary parts and real parts separately, taking into account that half of the values are just the negatives of the other half, reduces the effort by four. Furthermore, since the real and imaginary parts are only 2-bit values, multiplications may be avoided. 
         [0029]    In one envisaged application, the HSPA uplink, there are only three 16QAM symbols to be determined.  FIG. 3  illustrates how, in the HSPA uplink, two of the symbols (S 2 , S 3 ) use a length=2 CDMA code and the third (S 1 ) uses a length=4 code, and  FIG. 4  illustrates the corresponding symbol block structure. From  FIG. 4 , one sees that a succession of symbol blocks can be transmitted time-wise, with each symbol block including two length=2 code symbols (S 2 , S 3 ), and one length=4 code symbol (S 1 ). The symbols S 1 , S 2 , and S 3  within any one such block are efficiently detected according to the efficient multi-symbol detection method taught herein, by operating on the received signal samples representing the symbol block, where inter-block interference—i.e., interference arising from preceding and succeeding symbol blocks has been suppressed by other means, namely, a block-wise equalizer such as a DFE equalizer, inverse channel equalizer, or combination of the two, as described in the co-filed application to Bottomley identified herein below. Note, too, that the teachings presented herein apply directly to other symbol block structures. For example, in  FIG. 5 , one sees a stream of symbol blocks, each comprising three successive symbols, S 1 , S 2 , and S 3 . Of course, a lesser or greater number of symbols may be included in the symbol blocks of interest. 
         [0030]    Thus, as an example, a receiver according to one or more embodiments of the invention receives four chip-spaced signal samples (for a given symbol block) and may preprocess them to remove interference from previous or future symbols blocks. The resulting pre-processed values r 1 , r 2 , r 3 , r 4  may then be decoded according to one implementation of the invention as follows. 
         [0031]    Referring back, for example, to the  FIG. 3  symbol block structure and coding, the receiver forms R 2 =r 1 −r 2  (correlation with length=2 code  10 ); R 3 =r 3 −r 4  (correlation with length=2 code  10 ); and R 1 =r 1 +r 2 −r 3 −r 4  (correlation with length=4 code  1100 ). Separately, the receiver shall have determined channel coefficients which, together with the CDMA codes, and possibly other objects such as an overall complex scrambling code, are used to determine the 3×3 matrix of Q-values describing how the three symbols (S 1 , S 2 , and S 3 ) mix to produce the three correlations R 1 , R 2 , R 3 . 
         [0032]    More detail on preprocessing and on using channel estimates, CDMA codes and scrambling codes to compute the Q-matrix values are contained in U.S. application Ser. No. 12/035,846 (the &#39;846 application). As previously indicated, the instant application claims priority from and wholly incorporates by reference the &#39;846 application, which is commonly owned and co-pending with the instant application. In the &#39;846 application, block-based equalization is used to remove inter-block interference between symbol blocks in a stream of symbol blocks, and joint detection is used to detect the particular combination of symbols within a symbol block. Those block-based equalization teachings may be applied herein, as part of received signal sample preprocessing, such that the teachings provided herein for efficient multi-symbol detection may be understood as a particularly advantageous form of the joint detector taught in the &#39;846 application (e.g., as joint detector  82  shown in  FIG. 4  of that application.) 
         [0033]      FIG. 6  at least partly illustrates one embodiment of a receiver  10 , wherein a sample generator  12  produces pre-equalizer samples based on initially processing an (antenna-received) signal. These pre-equalizer samples may be chip samples, despread values, Rake-combined or Generalized Rake (G-Rake) combined values, etc. In any case, a block-equalizer  14 , which may be configured as a block-based decision feedback equalizer (DFE) as taught in the &#39;846 application, produces received signal samples that are pre-processed, at least in the sense that the signal samples representing a given symbol block have inter-block interference suppressed from them. Thus, the joint detector  16  is provided received signal samples for each symbol block of interest, for its use in efficiently performing multi-symbol detection (joint detection) of the symbols within each such symbol block of interest, according to the teachings presented herein. 
         [0034]      FIG. 7  provides more example details regarding the block-based DFE aspects of pre-processing. In particular,  FIG. 7  illustrates that the pre-equalizer sample values (obtained, e.g., by filtering, amplifying, downconverting, and despreading antenna-received signals), are input to the block equalizer  14 , which here is implemented as a block-based DFE  18 . The DFE  18  includes a combining circuit  20 , which performs subtractive cancellation of inter-block interference arising from preceding symbol blocks. It performs this function based on being supplied with appropriately filtered versions of prior detected values. These values are output by a feedback filter circuit  22  that has filter coefficients determined from prior-block dependence coefficients and is provided with prior detected symbols from the joint detector  16 . 
         [0035]    As shown, the output values from the combining circuit  20  serve as inputs to a feedforward filter circuit  24 , whose filter coefficients are determined according to succeeding block interference estimates, which may be based on impairment correlation estimates, etc., relating the effects of one or more succeeding symbol blocks on the sample values of a current symbol block of interest. The outputs of the feedforward filter circuit  24  thus are a set of received signal sample values that have been preprocessed for suppression of inter-block interference. Such sample values may be produced for each symbol block to be detected by the joint detector  16 . 
         [0036]    Continuing with the above joint detection example, then, the pre-processed values r 1 , r 2 , r 3 , r 4  or R 1 , R 2 , R 3 , R 4  for a given symbol block are decoded according to one implementation of the invention as follows. Once the Q-matrix values and the R-values are obtained, the symbol values S 1 , S 2 , S 3  are determined by the invention substantially as described above, except that only two symbols, S 2  and S 3 , need be hypothesized in order to determine the third, S 1 . Therefore, only 256 candidate symbol triples are produced for testing. The metrics for each of the 256 symbol triples are then computed efficiently, as described above, and the symbol triple yielding the lowest metric is the decoded result. 
         [0037]    If desired, once hard symbol decisions are available, they may be back substituted two at a time to give soft decisions for the third, by using Eq. (3), for example, and similar equations for the other symbols. Soft bit values can also be obtained by computing metric differences corresponding to symbols in which the bit of interest has been flipped. Soft decisions are useful when they are to be further processed by an error correction decoder, such as a turbo decoder. Turbo decoders generally process to refine bit log-likelihood ratios; therefore the step of converting soft symbol values to bitwise log-likelihood ratios generally precedes processing by a turbo decoder. 
         [0038]    The above method determines the most likely set of symbols from the received samples corresponding to the current multi-symbol, in the case that the noise on the different correlation values is uncorrelated and of equal RMS value, and ignoring interference from adjacent multi-symbols. In U.S. Pat. Nos. 6,963,532 and 6,215,762 to Applicant Dent, entitled “Communications system and method with orthogonal block encoding”, methods are described for removing adjacent multi-code symbol interference, therein termed “inter-vector interference” because the symbols of a multi-code symbol are considered to form a vector of symbols. The method comprises subtracting the effects of the earlier-determined adjacent multi-symbol, while using a feedforward filter with matrix coefficients to remove the effect of interference from as yet undetermined, future symbols. The &#39;532 and &#39;762 patents are also hereby incorporated by reference herein. Thus, techniques for suppressing inter-block interference in advance of sample processing for joint detection may be taken from the &#39;532 and &#39;762 patents, or from the earlier described &#39;846 application, which further describes how to determine the feedforward matrix coefficients better to account for other impairments such as noise and other signals. 
         [0039]    In the case where the noises on the R-values are not uncorrelated or are of unequal value, it is first necessary to modify Eq. (1) to obtain uncorrelated noises of equal RMS value. This may be done by multiplying both sides by a decorrelating matrix D, obtaining modified Q values and modified R values which may then be subjected to the inventive process as already described in order to decode the symbols. 
         [0040]    If (n) is a vector of the noises before decorrelation, then the noises become modified to (n′)=[D](n). The correlations between the (n′) are then given by the expected values of (n′)(n′) # =[D](n)(n) # [D] #  averaged over many noise samples. By substitution it is evident that [D](n)(n) # [D] # =I, the identity matrix, if [D] # [D]=(n)(n) # =[C] −1 , where C is the expected or average value of (n)(n)#, i.e., the correlation matrix. Because D is Hermitian, D # =D, and thus D 2 =C −1 . D is thus the square root of the inverse correlation matrix and is given by 
         [0000]        D=E   # Λ −1/2   E    
         [0000]    where E is the matrix of Eigenvectors of C and Λ is a diagonal matrix of its Eigenvalues. Thus letting Q′=DQ and R′=DR, we get 
         [0000]      [ Q ′]( S )=( R ′)  Eq. (6) 
         [0000]    Because Eq. (6) is of exactly the same form as Eq. (1), it may be solved for the best values of symbols S exactly as was already described for Eq. (1). 
         [0041]    This calculation of D may be based on averaging the noise correlation matrix over many symbol blocks, and thus does not necessarily need to be repeated for every symbol. However, it is preferable to determine the expected correlation matrix by expressing it in terms of the varying common scrambling codes combined with the less rapidly varying channel estimates. In this way, updated values of the correlation matrix are available for every symbol, due to the common scrambling codes changing in a known way, and it is desirable to determine the change to D thereby caused. Depending on the number of scrambling sequence chips on which the correlation matrix depends and the rate at which the channel changes, it may be more efficient to compute D for all combinations of those scrambling chips at intervals determined by the rate of change of channel coefficients. 
         [0042]    The issue of frequently recalculating the square-root matrix D may be avoided by processing received signal values r directly, before removal of the common, complex, scrambling code and before correlation with the Walsh codes.  FIG. 3  shows a particular example where three 16QAM symbols are coded into four CDMA chips r 1 , r 2 , r 3 , r 4 . The chips are linearly dependent on the symbol values, so that after reception through a multipath channel one may still write: 
         [0000]        Q 11· S 1+ Q 12· S 2+ Q 13· S 3= r 1, 
         [0000]        Q 21· S 1+ Q 22· S 2+ Q 23· S 3= r 2, 
         [0000]        Q 31· S 1+ Q 32· S 2+ Q 33· S 3= r 3, and 
         [0000]        Q 41· S 1+ Q 42· S 2+ Q 43· S 3= r 4. 
         [0043]    In the above, the values r 1 , r 2 , r 3 , r 4  may be received values that have already been preprocessed in some way, such as the use of block DFE to remove inter-symbol-vector interference from adjacent symbol vectors. The Q-values embody the complex scrambling code, the spreading codes assigned to each symbol, and multipath channel coefficients. In the absence of multipath, the Q-matrix would be given by 
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         [0000]    where X 1 , X 2 , X 3 , X 4  are the successive values of the common, complex, scrambling code applied to the four chips, and the three spreading codes assigned are (w 11 , w 21 , w 31 , w 41 ), (w 12 , w 22 , w 32 , w 42 ) and (w 13 , w 23 , w 33 , w 43 ). 
         [0044]    In the special example of  FIG. 3 , each chip depends on only two symbols, not three, and the Q-matrix, at least in the absence of multipath propagation is given by: 
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         [0000]    However, multipath propagation or preprocessing will likely have the effect of making the zero terms non-zero, so that in general all 4×3 Q-elements must be considered to be non-zero. The equations are now over-dimensioned, but may still be written as 
         [0000]      [ Q ]( S )=( r ). 
         [0000]    The noise correlation matrix and derived value of the decorrelating matrix D are now 4×4 matrices, giving decorrelated equations [Q′](S)=(r′) where Q′=DQ and r′=Dr, and now of size 3×4 and 1×4 respectively. By hypothesizing two of the symbols, such as S 2  and S 3 , subtracting their influence on r′ to obtain 
         [0000]        R 1′= r 1′− Q 12′· S 2− Q 13′· S 3, 
         [0000]        R 2′= r 2′− Q 22′· S 2− Q 23′· S 3, 
         [0000]        R 3′= r 3′− Q 32′· S 2− Q 33′· S 3, and 
         [0000]        R 4′= r 4′− Q 42′· S 2− Q 43′· S 3. 
         [0000]    The best value of the third symbol (e.g. S 1 ) is given by Eq. (3) by using the element values of Q′ instead of Q. That value of S 1  is then stored against the values of S 2  and S 3  used to obtain it, and metrics for each of the 256 symbol triples computed using Eq. (5) with the term in S 4  omitted. 
         [0045]    A further mathematical manipulation however avoids the need to compute square-root matrix D. First write the metric (R−QS) # C −1  (R−QS) in terms of the hypothesized symbols S′ # =(S 2 , S 3 , . . . ) and the non-hypothesized symbol S 1 . To do this, let Q # C −1 Q be denoted by the matrix [U], and partition U into its first column (u) and the rest, [Udim]. Then [Q](S)=(u)S 1 +[Udim](S′). Thus (R−QS)=(R−Udim·S′−(u)S 1 )=(R′−(u)S 1 ) where (R′)=(R)−[Udim](S′). Thus the metric becomes 
         [0000]        R′   #   C   −1   R− 2 Re{R′   #   C   −1   u·S 1+ u   #   C   −1   u·|S 1| 2 }. 
         [0000]    But u # C −1 u=a is a real scalar, and R′ # C −1 u=z* is a complex scalar. Therefore the soft value (unquantized value) of S 1  which minimizes the above metric is simply S 1 =z/a. The quantized symbol value is then the value in the symbol alphabet nearest to the soft value given above, the determination of which by separately quantizing real and imaginary parts to 2 bits was already explained above. Then, having obtained the quantized symbol S 1 , it is combined with the others to compute an associated metric using the above metric expression, which uses the matrix C −1  explicitly and does not require the square root of C −1  to be calculated. 
         [0046]    The method may also be used where there are more than four chips, each dependent on a set of symbols to be determined. For example, a diversity receiver having two antennas would supply four chips from each, and the resulting Q-matrix would be of dimension 8×3 for decoding three symbols, unless antenna signals are combined prior to detection. When chips from several antennas are processed to decode the symbols, use of the appropriate inverse correlation matrix C −1  has the additional benefit of reducing interference that has a different angle-of-arrival than the wanted signal. 
         [0047]    Other variations of the invention are possible. For example, in the special case of 16QAM symbols, it may be realized that the symbol value is expressible as a linear combination of its four constituent bits. Thus an equation similar to Eq. (1) can be written, equating received signal values or correlation values to a vector of bit values multiplied by a Q-matrix of dependency coefficients. 
         [0048]    In the method described above, we chose to hypothesize 8 bits constituting two symbols in order to obtain a candidate solution for the four bits of the third symbol. However, any 8 bits, or more or less, could in principle be hypothesized in order to obtain candidate values for the remainder, which would each be quantized to the ‘1’ value or the ‘0’ value. The candidate bit vectors would then be used to compute metrics and the bit vector having the best metric selected. 
         [0049]      FIG. 8  shows another embodiment of a receiver apparatus  10  for implementing the inventive multi-symbol detection processing taught herein. It should be understood that the illustrated embodiment may be simplified, e.g., transmission circuitry, user interface elements, and the like also may be present. Indeed, in at least one embodiment the receiver  10  comprises a wireless communication device, such as a cellular radiotelephone, PDA, pager, palmtop/laptop computer, network interface card, wireless adaptor/module, or other communication transceiver device or system. 
         [0050]    It also should be understood that  FIG. 8  may represent a functional circuit arrangement, rather than a literal physical circuit arrangement. For example, one or more of the illustrated entities may be implemented in a baseband signal processor or other digital processing circuit or circuits (e.g., microprocessor/DSP circuits). For example, the block DFE and joint detection processing functions described earlier regarding  FIG. 6 , and variations of them, can be incorporated into the processing elements illustrated in  FIG. 8 . 
         [0051]    Broadly, the functional processing of the illustrated entities may be implemented in hardware, software, or any combination thereof, and it should be understood that at least some entities can be implemented together in the same processing circuits, such as through appropriate software/firmware configuration. As such, at least part of the inventive processing taught herein may be implemented in the form of a computer program comprising program instructions for implementation on one or more digital processors. The computer program and any supporting data and configuration information can, as is well understood by those skilled in the art, be stored in a memory device included within the receiver  10 . 
         [0052]    With this implementation flexibility in mind, a signal sampler  100 - 1  is provided for initially processing signals received on an antenna (e.g., antenna  99 - 1 ). It filters, amplifies samples and converts the antenna-received signal to obtain a stream of digital sample values of at least one sample per symbol. As a non-limiting example, the receiver apparatus  10  is in one or more embodiments configured to obtain the received signal samples by processing a GSM/EDGE signal. In another non-limiting example, the receiver apparatus  10  is configured to obtain the received signal samples by processing an LTE signal. Also, it should be understood that the received signal samples may comprise samples of signals received using two or more antennas. 
         [0053]    In the case of multicode CDMA modulation, there are several chips per symbol and the signal may be sampled at several samples per chip in order to facilitate preprocessing, such as matched filtering, equalization of interference from adjacent multi-code symbols, or selection of an optimum sample timing. Optionally, additional signal samples  100 - 2  to  100 - n  can be used to sample signals from additional antennas (e.g.,  99 - 2 , . . . ,  99 - n ) for diversity reception, beamforming or spatial nulling, all of which are encompassed by proper determination of the aforementioned noise correlation matrix C, which is performed by optional noise correlation matrix estimator  220 . 
         [0054]    Estimation of the noise correlation matrix is beneficial when significant correlation between the noise plus interference on different signal samples is expected. Such is the case with using multiple antennas when interference arrives from distinct directions different than the wanted signal direction, or arrives through a multipath propagation channel with a different frequency response than the wanted signal propagation channel. Estimation of the noise correlation matrix may not be necessary in other applications. 
         [0055]    In any case, channel estimator  200  processes received signal samples to determine the multipath propagation channel for at least the wanted signal. Channel estimation can be done by correlating the received signal samples with a known code sequence that is included in the transmitted signal. If the known code sequence overlaps the information symbol bearing part of the wanted signal in both time and frequency, it may be subtracted out in processor  110  once the amount (amplitude, phase and delay) has been determined by channel estimator  200 . 
         [0056]    The above description of the invention includes correlating the received signal samples with CDMA codes and processing the correlations, or processing the received samples directly. Either can be done by processor  110  which is meant to be general enough to perform either. Likewise, noise correlation matrix estimator  220  is intended to be sufficiently general to estimate any or all of the noise correlation matrix, the inverse of the noise correlation matrix or the square root of the inverse of the noise correlation matrix, depending on the implementation chosen. Likewise, processor  110  is envisaged to compute dependence coefficients Q, Q′ based on the channel estimates and a priori knowledge of the symbol modulation and coding scheme, with the dependence coefficients describing how processed signal samples depend on the information symbols contained therein. 
         [0057]    Further, the processor  110  or another one of the receiver apparatus&#39;s one or more processing circuits may be configured to compute the first modified signal samples detailed herein by correlation of the received signal samples with a number of CDMA codes. Additionally or alternatively, the first modified signal samples also may be obtained by matrix multiplication of the received signal samples by the square root of the inverse of a noise correlation matrix. Similarly, the processor  110  or another one of the receiver apparatus&#39;s one or more processing circuits may be configured to compute the modified dependence coefficients Q′ based on matrix multiplication by the inverse of a noise correlation matrix. 
         [0058]    In any case, symbol sub-group generator  140  generates, in turn, all possibilities for a subgroup of the information symbols carried by the received signal samples. For example, the subgroup can comprise all except one 16QAM symbol. The subgroup is presented to processor  110  which subtracts its influence using the channel estimates from channel estimator  200 , a priori knowledge of the signal modulation and coding (or the dependence coefficients derived therefrom) and the appropriate derivative of the noise correlation matrix, if used. Remaining soft symbol decoder  120  determines the remaining symbols in the set which are not hypothesized by symbol subgroup generator  140 , by combining processed samples from processor  110 , again using the appropriate derivative of the noise correlation matrix, if required. Soft symbols from soft symbol decoder  120  are then quantized by symbol quantizer  130  to the nearest symbol value in the symbol alphabet. The quantized symbol from quantizer  130  together with the subgroup from subgroup generator  140  then form a complete set of symbols which is a candidate symbol set. 
         [0059]    A candidate symbol set is produced for each symbol subgroup generated by subgroup generator  140  and presented to metric computer  150  which computes a metric based on processed samples from processor  110 , the candidate symbol set, and the appropriate derivative of the noise correlation matrix if required. The metric computed describes how well the candidate symbol set explains the received samples, given the channel estimates and the noise correlation matrix. Candidate symbol set selector  160  then selects the candidate symbol set that yields the best metric. 
         [0060]    Thus, the one or more processing circuits of the receiver apparatus  10  (e.g., processor  110 , decoder  120 , quantizer  130 , subgroup generator  140 , metric computer  150 , and candidate set/best metric selector  160 ) implement an efficient method of joint detection, where multiple symbols within a symbol block of interest are efficiently decoded according to the teachings presented herein. For example, using the inventive apparatus of  FIG. 8  to process received signals such as shown in  FIG. 3 , which comprises three 16QAM symbols, only 256 metrics have to be computed to determine the most likely set of three symbols. This processing is a significant improvement over conventional “brute force” methods which would require computation of 4096 metrics for determination of the three 16QAM symbols. (It also should be understood that these processing circuits, e.g., processor  110 , or other processing elements within the receiver apparatus  10 , can be configured to provide received signal sample preprocessing. Such processing preferably includes, as discussed for  FIGS. 6 and 7 , block-based DFE equalization of the received signal samples for a given symbol block, to remove the effects of inter-block interference.) 
         [0061]      FIG. 9  illustrates one embodiment of processing efficient multi-symbols detection, where the receiver apparatus  10  of  FIG. 8  can be configured via hardware or software to carry out the illustrated processing operations. Regardless of the hardware and/or software implementation details,  FIG. 9  illustrates one embodiment of a method for decoding a group of symbols from a set of received signal samples that depend on each symbol of the group to an extent depending on an associated dependence coefficient. For example, using earlier-introduced notation, the receiver apparatus  10  may obtain received signal samples r 1 , r 2 , r 3 , and r 4 , corresponding to a symbol block of interest having symbols S 1  (length=4 code), and symbols S 2  and S 3  (length=2 code). 
         [0062]    The illustrated method includes computing first modified signal samples and modified dependence coefficients using said dependence coefficients and said received signal samples (Block  250 ). For example, the first modified signal samples are formed as the R values described earlier herein, and the modified dependence coefficients are formed as the Q′ values obtained from the Q values. 
         [0063]    Processing continues with, for each possible combination of all except at least one symbol in the group of symbols, subtracting from said first modified signal samples the dependence on said combination of symbols as described by said modified dependence coefficients in order to obtain second modified signal samples dependent principally on the remaining at least one symbol (Block  252 ). As one example of obtaining second modified signal samples, one may refer to the processing represented in Eq. (2). 
         [0064]    Processing continues with determining unquantized values for said remaining at least one symbol from said second modified signal samples (Block  254 ). See, for example, the discussion immediately below Eq. (3) for an illustration of an unquantized value derived in accordance with one embodiment of this processing operation. Correspondingly, processing continues with quantizing each of the unquantized remaining at least one symbol to the nearest symbol value in the symbol alphabet to obtain quantized values for said at least one remaining symbol, and associating the quantized symbol values with the combination of other symbols to obtain a candidate symbol group (Block  256 ). It also is possible to go directly from the second modified samples to quantized values. Thus, it should be understood that the illustrated processing may be modified to go from the second modified values to the quantized values, without necessarily obtaining unquantized values. 
         [0065]    Still further, the illustrated processing continues with, for each candidate symbol group, computing a metric that describes how well the candidate symbol group explains said set of received signal samples, and selecting the candidate group giving the best metric (Block  258 ). Here, selecting the candidate group giving the best metric represents the decoded symbols for the at least one remaining symbol. It should be understood that the other symbols in the group can be decoded by repeating these processing operations with those other symbols individually or collectively being considered “the at least one remaining symbol.” 
         [0066]    Thus, the teachings herein present a method for decoding a group of symbols from a set of received signal samples that depend on each symbol of the group to an extent depending on an associated dependence coefficient. In at least one embodiment, the method comprises, for each selected subset of symbols to be decoded from a group of symbols, forming hypothesized subsets representing different combinations of possible symbol values for those remaining symbols in the group not included in the selected subset. The method continues with, for each hypothesized subset, generating an estimate of the symbols in the selected subset. The method further continues with forming candidate symbol sets from the hypothesized subsets and the corresponding estimates. Each candidate symbol set thus represents a different combination of possible symbol values for the group of symbols. The method further continues with computing a metric for each candidate symbol set, the value of which indicates how well the candidate symbol set corresponds to the received signal samples, and outputting as decoded symbols the estimate of the symbols in the selected subset that corresponds to the candidate symbol set having a best one of the metrics. Here, “best” indicates closest matching or the like, and may be detected from the largest or smallest metric (e.g., magnitude evaluation) depending on whether the metric represents a maximization or minimization function. 
         [0067]    Yet another formulation of the invention will now be presented in which equations are formulated with all-real quantities. It was already shown above how one optimum solution in the face of correlated noise is first to modify the equation [Q](S)=R by multiplying both sides by [D], the square root of the inverse correlation matrix to obtain 
         [0000]      [ Q ′]( S )=( R ′)  Eq. (7) 
         [0000]    where [Q′]=[D][Q] and (R′)=[D](R). In Eq. (7), all quantities are complex, but when the symbols are for example M-ary QAM symbols, they can be partitioned into real and imaginary parts that each depend on a subset of the constituent bits. This is not necessarily true for all modulations, or even all M-QAM constellations, but is true at least when M is an even power of two, such as 16 or 64. We can thus define A 1  to be the real, 2-bit part of complex symbol S 1  and A 2  to be the imaginary 2-bit part. Likewise let A 3 +jA 4 =S 2 , A 5 +jA 6 =S 3 , and so forth. 
         [0068]    In terms of the 2-bit A&#39;s, Eq. (7) (for the 3×3 case) can be written as 
         [0000]    
       
         
           
             
               
                 
                   
                     
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         [0000]    where QR, QI refer to real and imaginary parts respectively of a Q-element, and RR, RI refer respectively to real and imaginary parts of the signal values R. It shall also be understood that the Q and R values are the primed values if the option of pre-multiplying by the decorrelation matrix D has been employed, or not primed otherwise. 
         [0069]    Now Eq. (8) represents six equations for the same number of variables, but as already stated, a direct solution by matrix inversion is not necessarily a good solution. In fact, the direct solution erases the effect of multiplying both sides by D, so no benefit of noise decorrelation would be obtained in that case. However, if we hypothesize just one of the A&#39;s, and remove its effect to the RHS, we are left with six equations in just five remaining A-&#39;s, which is now an over-dimensioned problem, to which a least squares solution exists. Once that least squares solution has been found, the resulting “soft” A-values are quantized to the nearest 2-bit value, and associated with the originally hypothesized A-value. The entire set of 12 bits is then stored as one candidate solution. Four candidates are produced by this means, for all values of the originally hypothesized A-value, and then tested to see which best fulfills Eq. (7). It can now be seen that any number n of the A-values can be hypothesized, n=1 to 5, and the remaining 6-n are then found by quantizing the least squares solution for the remaining A-values. 
         [0070]    When n A-values are hypothesized and their contributions removed to the RHS of Eq. (8), the remaining matrix is reduced in dimension to 6×(6−n). Denoting this diminished matrix by [Qdim] and the modified RHS by (R″) and the remaining symbols by (Adim), we have to solve 
         [0000]      [ Qdim ]( Adim )=( R ″),  Eq. (9) 
         [0000]    for the least squares solution. By standard linear algebra, the solution is known to be: (Adim)=[Q T dim·Qdim] −1 [Q T ] (R″). This solution to Adim is then quantized to two bits element by element and stored together with the n originally hypothesized A values as a candidate solution, and a metric computed for it describing how well it satisfies original Eq. (8). The candidate having the best metric, i.e., the lowest residual sum-square error in satisfying the equations represented in Eq. (8), is then selected as the set of 12 decoded bits. 
         [0071]    It can be mentioned that, in a multicode system in which not all codes are allocated to a receiver, and one code remains idle or is modulated with known data such as a pilot code used for channel estimation, it can still be useful to perform and utilize correlations with the unused code. Although the matrix of dependence coefficients Q will contain zeros in the equation describing dependence on the wanted symbols of the correlation with the extra code or codes, these zeros become non-zero values when the matrix is multiplied by the decorrelating matrix D. Thus it may still be useful to utilize signal samples or correlations that give no clue about the wanted symbols, because they instead may give a clue about the noise that affects the wanted symbols. 
         [0072]    The above formulation may be able to be used for other modulations, providing their real and imaginary parts can be described in terms of a linear combination of bits with suitable coefficients. In finding such a description, the assignment of bit patterns to constellation points may be chosen arbitrarily for the purposes of deciding which symbols were received, as long as the symbols thereafter are assigned the true bit patterns as intended by the transmitter. 
         [0073]    We have thus shown how the effort in decoding a multicode symbol, or other type of symbol block of two or more symbols, may be reduced by at least a factor equal to the size of the symbol alphabet, which would be an even greater saving with larger alphabets such as 64QAM. The equations may be arranged and computed in various ways by a person skilled in the art. As such, it should be understood that the foregoing description and the accompanying drawings represent non-limiting examples of the teachings presented herein and do not limit the present invention. Instead, the present invention is limited only by the following claims and their legal equivalents.