Abstract:
A dual band antenna for a mobile communication system which includes: a metal tube having an open end; a coaxial feed line having inner and outer conductors, with one portion of the coaxial line inserted into the metal tube, a ground plane connected to a portion of the metal tube opposite the open end and to the outer conductor of the coaxial feed line, and, a signal line electrically coupled to the inner conductor and protruding from the metal tube at the open end thereof. Preferably, the dimensions of the metal tube, the coaxial feed line and the signal line are selected such that the antenna is impedance matched to the coaxial feed line over the dual operating band, thereby obviating the need for a separate matching network.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to an antenna and, more particularly, to a dual band antenna in which a separate matching circuit is not required between a signal source and the antenna, thereby having simple construction, convenient usage, low price and enhanced performance. 
     2. Description of the Related Art 
     In a mobile communication system, an antenna serves to conserve transmitting power and to use frequency efficiently. With the rapid development and widespread usage of mobile communications, there are frequent occurrences of capacity saturation in a conventional system. Thus, there is a need for a new system which works well in such environment and an interlock between a conventional system and the new system. For example, interlocks are used between: (i) a Code Division Multiple Access (CDMA) system and a Personal Communication System (PCS) in Korea; (ii) an Advanced Mobile Phone Service (AMPS) system and a PCS in the United States; (iii) a Groupe Special Mobile (GSM) system and a Digital European Cordless Telephone (DECT) system; or (IV) a GSM system and a Digital Communication System (DCS) 1800 system, applying the GSM to band 1,800 MHz in Europe. Such interlock systems are commonly called dual band systems. That is, a dual band system interlocks two different systems having frequency bands different from each other. 
     In conventional dual-band systems having different antennas for the respective two bands, there exists duplication in material costs which makes miniaturization and weight reduction difficult. Therefore, a dual band antenna usable at two bands has been developed. 
     U.S. Pat. No. 4,509,056 discloses a multi-frequency antenna employing tuned sleeve chokes. FIG. 1 is a section view illustrating the construction of a monopole antenna operating at dual frequency according to an embodiment of the multi-frequency antenna employing tuned sleeve chokes. As shown in FIG. 1, an outer conductor  6  of a coaxial feed line  2  is connected to a ground plane  20  and an extension  10  of an inner conductor  8  is extended from the ground plane  20  passing through a choke  12   i  to a radiating section indicated as dimension N. The choke is loaded with a solid dielectric insert  16   i  and the inner surface of the shell of the choke and the outer surface of the conductor extending through the choke form a quarter wavelength (λ/4) transmission line. At high frequency, the choke forms a λ/4 transmission line which prevents coupling between an open end of shell  14   i  of the choke  12   i  and the extension  10 . At low frequency, the choke  12   i  is operated not as an isolation element but as a monopole antenna indicated as the entire length P at the low resonant frequency. 
     The dual band antenna operating as a quarter wavelength monopole antenna at high/low band frequencies has an input impedance Z in  as defined in equation 1 and requires a 50Ω matching circuit in the case where it is connected to another circuit of the system. Here, the other circuit means a filter or a radio frequency (RF) amplifier, and when it is connected to the dual band antenna, the performance of the antenna is reduced due to impedance mismatching. Therefore, the 50Ω matching circuit should be connected for preventing the mismatching as described above. 
     
       
           Z   in =36 +j 20  (1) 
       
     
     Since the above dual band antenna requires a separate matching circuit between a signal source and the antenna, it results in complicated construction, inconvenience of usage, and high price. 
     SUMMARY OF THE INVENTION 
     It is therefore an object of the present invention to provide a dual band antenna in which a separate matching circuit is not required between a signal source and the antenna, thereby having simple construction, convenient usage, low price and enhanced performance. 
     To achieve the above objects, an embodiment of the present invention is provided, that is, a dual band antenna for a mobile communication system which includes: a metal tube having an open end; a coaxial feed line having inner and outer conductors, with one portion of the coaxial line inserted into the metal tube. A ground plane is connected to a portion of the metal tube opposite the open end and to the outer conductor of the coaxial feed line. A signal line is electrically coupled to the inner conductor and protrudes from the metal tube at the open end thereof. 
     Preferably, the dimensions of the metal tube, the signal line, and the coaxial line are selected such that the impedance of the antenna is substantially matched to the impedance of the coaxial feed line over the dual band of operation. Optionally, the metal tube can be filled with dielectric to shorten the antenna length. 
     In a more specific embodiment, a dual band antenna for a mobile communications system includes a metal tube, a coaxial feed line having one portion inserted into the metal tube, a ground plane connected to a first end of the metal tube and to the outer conductor of the coaxial feed line, and a signal line. The signal line is connected to the inner conductor of the coaxial line at a connection point within the metal tube. The outer diameter of the coaxial line is open at the connection point, thus creating a first radio frequency (RF) choke. The metal tube has a second end that is open to create a second RF choke. The signal line passes through the metal tube and protrudes past the metal tube by a predetermined length. Predetermined values are established for: the length of the coaxial line from the ground plane to the connection point; the length of the signal line from connection point to the open end of the metal tube; the length of the protruded signal line from the open end of the metal tube; the outer diameter of the outer conductor of the coaxial feed line; and diameters of the metal tube and the signal line. These values are selected such that, in an operating frequency band of said antenna, impedance of the antenna substantially matches impedance of the coaxial line, thereby providing a low standing wave ratio on the coaxial line, and obviating the need for a separate matching network. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     These and various other features and advantages of the present invention will be readily understood with reference to the following detailed description taken in conjunction with the accompanying drawings, wherein: 
     FIG. 1 is a section view illustrating the construction of a prior art monopole antenna operating at dual frequency having tuned sleeve chokes; 
     FIG. 2 is a section view illustrating the construction of a dual band antenna according to an embodiment of the present invention; 
     FIG. 3 is a diagram illustrating an equivalent circuit of the dual band antenna shown in FIG. 2; 
     FIG. 4 is a diagram illustrating an equivalent circuit of FIG. 2, once-simplified by combining signal source and impedance Z AB ; 
     FIG. 5 is a diagram illustrating an equivalent circuit of FIG. 2, twice-simplified; 
     FIG. 6 is a diagram illustrating an equivalent circuit of FIG. 2, thrice-simplified by combining impedance Z** and impedance Z CD  to form impedance Z EF  viewed from points E and F; 
     FIG. 7 is a diagram illustrating the equivalent circuit in which dielectric constant, d 1 , d 2 , l 1 , l 2  and l 3  have predetermined values embodying the dual band antenna according to an embodiment of the invention; 
     FIG. 8 is a diagram illustrating a radiation pattern measured in comparison with a standard dipole antenna and the dual band antenna according to an embodiment of the present invention; 
     FIG. 9 is a diagram illustrating impedance characteristic of the dual band antenna according to an embodiment of the present invention; 
     FIG. 10 is a diagram illustrating standing-wave ratio (SWR) of the dual band antenna according to an embodiment of the present invention; and 
     FIG. 11 illustrates another embodiment of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     Hereinafter, a preferred embodiment of the present invention will be described in detail with reference to the accompanying drawings. Throughout the drawings, the same reference numerals or letters will be used to designate like or equivalent elements having the same function. Furthermore, in the following description, numerous specific details such as preferred components composing the circuit are set forth to provide a more thorough understanding of the present invention. It will be apparent, however, to one skilled in the art that the present invention may be practiced without these specific details. Known function and construction unnecessarily obscuring the subject matter of the present invention will be avoided in the detailed description of the present invention. 
     FIG. 2 is a section view illustrating the construction of the dual band antenna according to the embodiment of the present invention, which consists of a coaxial feed line  30 , a choke  60  comprising a metal tube  40  and a dielectric material  80 , a signal line  15  and a ground plane  50 . Herein, reference marks A to B are only used for understanding of the relation between FIG.  2  and the associated drawings showing equivalent circuits. 
     Preferably, one end of the metal tube  40  is connected to the ground plane  50 , and the other end thereof is open. The physical length of the metal tube  40  is approximately one quarter wavelength (l 1 +l 2 ) at the central frequency of a high frequency band. The coaxial feed line  30  is comprised of an inner conductor  70  and an outer conductor  25 , wherein one portion thereof is inserted into the metal tube  40 . The outer conductor  25  of the coaxial feed line  30  is connected to the ground plane  50 . The above portion of the coaxial feed line  30  inserted into the metal tube is extended from the ground plane  50  toward the opened end of the metal tube  40  by the length indicated as l 1 . The inner conductor  70  is connected (at point K) to a signal line  15  having the same diameter d 3  as the diameter d 1  of the outer conductor  25  of the coaxial feed line  30  at the end of the coaxial feed line  30  inserted into the metal tube  40 . At point K, the outer conductor  25  of coaxial line  30  is open, thus creating an RF choke (i.e., coaxial line  30  ends at point K). The opposite end of the coaxial line is connectable to electronics (not shown), used in conjunction with the antenna, such as a transmitter and/or receiver. The signal line  15  is passed through the open end of the metal tube  40 , but protrudes past the open end of the metal tube  40 . The metal tube  40  has the diameter d 2  and is filled with dielectric material  80 . Since the dielectric material has a dielectric constant higher than that of air, it allows the length of the metal tube  40  to be shorter for a given electrical length (as compared to an air-filled metal tube). The open end of metal tube  40  creates a second RF choke. 
     FIG. 3 is a diagram illustrating an equivalent circuit of the dual band antenna shown in FIG.  2 . An operation of the equivalent circuit of the dual band antenna will be described hereafter. 
     Z AB  represents the impedance of the choke  60  which comprises the metal tube  40  having its width from points A to B, the coaxial feed line  30  and the dielectric material  80  filling the metal tube  40 . Z AB  is represented by equation (2) since it is theoretically operated as a short-line.                  Z   AB     =       Zo                   tanh        (     γ                   l   1       )                       Z   0       =       60       ɛ   R            ln          d   2       d   1             ,           (   2   )                 γ   =     α   +     j                 k         ,     K   =       2      π     λ                                                
     Excluding the attenuation constant α from equation (2) results in equation (3).                Z   AB     =         jZ   0          tan        (     K                   l   1       )         =     j                   60       ɛ   R            ln          d   2       d   1            tan        (     K                   l   1       )                   (   3   )                                
     wherein, 
     α: damping element, 
     K: propagation constant, 
     Z 0 : characteristic impedance of short-line, 
     l 1 : length from ground plane to open end of coaxial feed line, 
     d 1 : outer diameter of outer conductor of coaxial feed line, 
     d 2 : inner diameter of metal tube, 
     λ: wavelength, and 
     ∈ R : relative dielectric constant of dielectric material. 
     The equivalent impedance Z CD  is calculated by the above equations 2 and 3, and if the diameter d 1  of the outer conductor  25  of the coaxial feed line  30  is equal to the diameter d 3  of the signal line  15 , the impedance Z CD  can be calculated by changing length (l 1 +l 2 ). The impedance Z CD  is indicated by equation (4).                Z   CD     =     j                   60       ɛ   R            ln          d   2       d   1            tan        [         2      π     λ          (       l   1     +     l   2       )       ]                 (   4   )                                
     FIG. 4 is a diagram illustrating an equivalent circuit once-simplified by combining signal source and impedance Z AB . Impedance Z* is represented by equation (5).                Z   *     =       50   ·     Z   AB           Z   AB     +   50               (   5   )                                
     FIG. 5 is a diagram illustrating an equivalent circuit twice-simplified with the simplified equivalent circuit shown in FIG.  4 . Referring to FIG. 2, since the length l 2  from the upper end of coaxial feed line  30  to the open end of the metal tube  40  is constructed and operated as one portion of the signal line  15  and the metal tube  40 , if it is combined with the impedance Z*, the equivalent circuit as shown in FIG. 4 can be obtained. An impedance Z** according to the equivalent circuit of the FIG. 4 can be obtained as indicated by equation (6).                  Z   **     =         Z   0                       [         Z   *       Z   0       +     tanh        (     γ                   l   2       )         ]       [   1   +         Z   *       Z   0          j                   tanh   (     γ                   l   2       ]             =       [         Z   *       Z   0       +     j                   tan        (         2      π     λ          l   2       )           ]       [     1   +         Z   *       Z   0          j                     tan   (         2      π     λ          l   2         ]                 ,           (   6   )                                
     FIG. 6 is a diagram illustrating an impedance Z EF  viewed from points E and F in an equivalent circuit in which impedance Z** and impedance Z CD  are combined. Then, impedance Z EF  can be obtained as indicated by equation (7).                Z   EF     =         Z   **          Z   CD           Z   **     +     Z   CD                 (   7   )                                
     Accordingly, impedance Z EF  is calculated by changing variables such as frequency, dielectric constant, d 1 , d 2 , l 1 , l 2  and l 3 . 
     FIG. 7 is a diagram illustrating the equivalent circuit in which dielectric constant, d 1 , d 2 , l 1 , l 2  and l 3  have predetermined values embodying a dual band antenna according to an embodiment of the present invention. Since signal source impedance Z EF  varies with operating frequency, it will be designated as Z EF (f). Z EF (f) is shown in FIG. 6 having the antenna as a load. Since antenna impedance Z ANT  also varies with frequency, it will be designated as Z ANT (f). Accordingly, signal source impedance Z EF (f), having an integral variable matching circuit, thus equals antenna impedance Z ANT (f). Therefore, in the embodiment of the present invention, dielectric constant variables, d 1 , d 2 , l 1 , l 2  and l 3  are varied upon construction of impedance Z EF (f), so that impedance Z EF (f) and impedance Z ANT (f) can be embodied to be equal to each other. Thus, a matching condition between the signal source and the antenna can be exactly achieved and can improve the characteristics of the dual band antenna. 
     FIG. 8 is a diagram illustrating a radiation pattern measured in comparison with a standard dipole antenna and the dual band antenna according to an embodiment of the present invention. FIG. 9 is a diagram illustrating impedance characteristic of the dual band antenna according to an embodiment of the present invention, and FIG. 10 is a diagram illustrating a standing-wave ratio (SWR) of the dual band antenna according to an embodiment of the present invention. At this moment, the CDMA and Korean PCS frequency of the dual band antenna will be given as follows: the CDMA frequency is 824˜849 MH z  upon transmission and 869˜894 MH z  upon reception; the Korean PCS frequency is 1750˜1780 MH z  upon transmission and 1840˜1870 MH z  upon reception. Because the dual band antenna may be applied to systems of the GSM/DECT, GSM/DCS 1800, the AMPS and CDMA/PCS, it can be easily made by varying first length l 1  and second length l 2  of the choke  60  divided at the point (point K) in which the inner conductor  70  of the coaxial feed line  30  and the signal line  15  are connected with each other as shown in FIG.  2 . If the length l 1 +l 2  of the choke  60  is varied, the resonant point of the high frequency band is moved, however, as indicated in FIG. 10 by the dotted line  81 , the resonant point of the low frequency band is barely moved as shown in FIG.  10 . Referring to the solid line as thickly indicated in FIG. 9, an interval Δ from a start point to point  3 , covering points  1  and  2 , shows the characteristic of the low frequency band (824˜894 MH z ). An interval that returns to the start point by covering points  3  and  4  indicated as Δ shows the characteristic of the high frequency band (1,750˜1,870 MH z ). The intervals between points  1  and  2 , between  3  and  4 , are the same as those as shown in FIG.  10 . 
     FIG. 11 shows an alternate embodiment of the present invention. This embodiment is similar to the embodiment described in connection with FIG. 2, except that the signal line  15  is replaced by a signal line  15 ′. Signal line  15 ′ consists of a first linear portion  15   a  of length l 2  and a spiral portion  15   b  of length l 3 . The diameter of both portions  15   a  and  15   b  is d 3 , although the diameter of the spiral portion  15   b  may be selected different than the linear portion  15   a.    
     The embodiments of the present invention described above have advantages in that a variable matching circuit is provided, thus a separate matching circuit is not required between a signal source and an antenna, thereby having a simple construction, convenient usage, low price and enhanced performance. 
     While what has been illustrated and described is considered to be the preferred embodiments of the present invention, it will be understood by those skilled in that art that various changes and modifications may be made, and equivalents may be substituted for elements thereof, without departing from the true scope of the present invention.