Abstract:
A current switching cell for a digital to analog converter. The switching cell includes three stages, a first control stage, a data stage, and a second control stage. The first control stage is configured to either disconnect the outputs of the digital to analog converter, or to connect them to the outputs of the data stage. The data stage is configured to operate in one of two states, depending on a data signal received, and the second control stage is configured to selectively invert the output of the digital to analog converter. The two control stages may be driven with several combinations of control waveforms to implement a non return to zero mode, a return to zero mode, inverse non return to zero mode, and inverse return to zero mode.

Description:
GOVERNMENT LICENSE RIGHTS 
     This invention was made with Government support. The Government has certain rights in the invention. 
    
    
     BACKGROUND 
     1. Field 
     One or more aspects of embodiments according to the present invention relate to digital to analog conversion, and more particularly to a system and method for generating wideband output signal from a digital to analog converter (DAC). 
     2. Description of Related Art 
     In applications using a digital to analog converter and requiring synthesized tones at frequencies that may be well above the sampling rate of the digital to analog converter, it may be desirable to use tones in the second Nyquist zone or in higher Nyquist zones. The power in such tones is attenuated by the sinc function of a conventional zeroth order hold DAC response beyond the first Nyquist zone and may therefore be too low to meet some requirements. 
     Thus, there is a need for a system and method for generating relatively high-power tones, from a digital to analog converter, in Nyquist zones above the first Nyquist zone. 
     SUMMARY 
     According to an embodiment of the present invention there is provided a digital to analog converter including: a current switching cell, the cell including: a first control stage having a first control input, first and second current inputs, and first and second current outputs, the first control stage being configured to: connect each current output of the first control stage to a respective current input of the first control stage when the first control input is in a first state, and disconnect each current output of the first control stage when the first control input is in a second state; a data stage having a data input, first and second current inputs, and first and second current outputs connected to respective current inputs of the first control stage, the data stage being configured to: connect the first and second current outputs of the data stage to the first and second current inputs of the data stage, respectively, when the data input is in a first state and, connect the first and second current outputs of the data stage to the second and first current inputs of the data stage, respectively, when the data input is in a second state; and a second control stage, having a second control input, a current input and first and second current outputs connected to respective current inputs of the data stage, the second control stage being configured to: connect the first current output of the second control stage to the current input of the second control stage when the second control input is in a first state, and connect the second current output of the second control stage to the current input of the second control stage when the second control input is in a second state. 
     In one embodiment, the first control stage includes a first transistor connected between the first current input of the first control stage and the first current output of the first control stage, and a second transistor connected between the second current input of the first control stage and the second current output of the first control stage. 
     In one embodiment, the first transistor is an n-channel metal-oxide semiconductor field effect transistor, a source of the first transistor being connected to the first current input of the first control stage and a drain of the first transistor being connected to the first current output of the first control stage. 
     In one embodiment, the data stage includes: a first differential pair having a tail, a first current output, and a second current output; and a second differential pair having a tail, a first current output, and a second current output, the first current output of the first differential pair and the first current output of the second differential pair being connected to the first current output of the data stage, the second current output of the first differential pair and the second current output of the second differential pair being connected to the second current output of the data stage, and the inputs of the first and second differential pairs being cross-coupled. 
     In one embodiment, the second control stage includes a differential pair. 
     In one embodiment, the digital to analog converter includes a resistor network connected to the current outputs of the first control stage. 
     In one embodiment, the digital to analog converter includes: a plurality of additional current switching cells; and a plurality of current references, each current reference being configured to source or sink a current having a first magnitude, each current reference being connected to a respective one of the current switching cell and the additional current switching cells. 
     In one embodiment, the digital to analog converter includes a resistor network connected to the current outputs of the first control stage, wherein the current switching cell and the additional current switching cell are connected to the resistor network. 
     In one embodiment, the resistor network includes: a first termination resistor connected between the first current output of the two current outputs of the first control stage and AC ground; and a second termination resistor connected between the second current output of the two current outputs of the first control stage and AC ground. 
     In one embodiment, the resistor network includes an R-2R ladder network. 
     In one embodiment, the digital to analog converter includes a control circuit having a clock input for receiving a periodic clock signal, a first control output connected to the first control input, and a second control output connected to the second control input, the control circuit being configured to operate in a first mode in which the control circuit: sets the first control input to the first state, and sets the second control input to the first state; the control circuit being further configured to operate in a second mode in which the control circuit: sets the first control input to the first state during one half of each cycle of the clock signal and to the second state during the remainder of each cycle of the clock signal, and sets the second control input to the first state; the control circuit being further configured to operate in a third mode in which the control circuit: sets the first control input to the first state, and sets the second control input to the first state during one half of each cycle of the clock signal and to the second state during the remainder of each cycle of the clock signal; and the control circuit being further configured to operate in a fourth mode in which the control circuit: sets the first control input to the first state during a first portion of each cycle of the clock signal and to the second state during the remainder of each cycle of the clock signal, and sets the second control input to the first state during a first portion of the first portion of each cycle of the clock signal and to the second state during the remainder of the first portion of each cycle of the clock signal. 
     In one embodiment, the first portion of each cycle of the clock signal is the first one-half of each cycle of the clock signal. 
     In one embodiment, the first portion of the first portion of each cycle of the clock signal is the first one-half of the first portion of each cycle of the clock signal. 
     In one embodiment, the first portion of each cycle of the clock signal is the first two-thirds of each cycle of the clock signal. 
     In one embodiment, the first portion of the first portion of each cycle of the clock signal is the first one-half of the first portion of each cycle of the clock signal. 
     In one embodiment, the first control stage includes a first transistor connected between the first current input of the first control stage and the first current output of the first control stage, and a second transistor connected between the second current input of the first control stage and the second current output of the first control stage. 
     In one embodiment, the data stage includes: a first differential pair having a tail, a first current output, and a second current output; and a second differential pair having a tail, a first current output, and a second current output, the first current output of the first differential pair and the first current output of the second differential pair being connected to the first current output of the data stage, the second current output of the first differential pair and the second current output of the second differential pair being connected to the second current output of the data stage, and the inputs of the first and second differential pairs being cross-coupled. 
     In one embodiment, the digital to analog converter includes: a plurality of additional current switching cells; and a plurality of current references, each current reference being configured to source or sink a current having a first magnitude, each current reference being connected to a respective one of the current switching cell and the additional current switching cells. 
     In one embodiment, the digital to analog converter includes a resistor network connected to the current outputs of the first control stage, wherein the current switching cell and the additional current switching cell are connected to the resistor network. 
     In one embodiment, the resistor network includes: a first termination resistor connected between the first current output of the two current outputs of the first control stage and AC ground; and a second termination resistor connected between the second current output of the two current outputs of the first control stage and AC ground. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Features, aspects, and embodiments are described in conjunction with the attached drawings, in which: 
         FIG. 1  is a waveform diagram showing four modes of operation of a digital to analog converter, according to an embodiment of the present invention; 
         FIG. 2  is a graph of spectral envelopes for four modes of operation of a digital to analog converter according to an embodiment of the present invention; 
         FIG. 3  is a schematic diagram of a switching cell for a digital to analog converter according to an embodiment of the present invention; 
         FIG. 4  is a block diagram of a digital to analog converter according to an embodiment of the present invention; 
         FIG. 5  is a waveform diagram showing a first mode of operation of a digital to analog converter, according to an embodiment of the present invention; 
         FIG. 6  is a waveform diagram showing a second mode of operation of a digital to analog converter, according to an embodiment of the present invention; 
         FIG. 7  is a waveform diagram showing a third mode of operation of a digital to analog converter, according to an embodiment of the present invention; 
         FIG. 8  is a waveform diagram showing a fourth mode of operation of a digital to analog converter, according to an embodiment of the present invention; 
         FIG. 9  is a waveform diagram showing a variant of the fourth mode of operation of  FIG. 8 , according to an embodiment of the present invention; and 
         FIG. 10  is a schematic diagram of a switching cell for a digital to analog converter according to an embodiment of the present invention. 
     
    
    
     DETAILED DESCRIPTION 
     The detailed description set forth below in connection with the appended drawings is intended as a description of exemplary embodiments of a wideband multi-mode current switch for a digital to analog converter provided in accordance with the present invention and is not intended to represent the only forms in which the present invention may be constructed or utilized. The description sets forth the features of the present invention in connection with the illustrated embodiments. It is to be understood, however, that the same or equivalent functions and structures may be accomplished by different embodiments that are also intended to be encompassed within the spirit and scope of the invention. As denoted elsewhere herein, like element numbers are intended to indicate like elements or features. 
     Referring to  FIG. 1 , in one embodiment a digital to analog converter (DAC) is configured to generate an analog signal at an analog output in a non return to zero (NRZ) mode. In this mode the analog output is updated once every clock cycle, on the rising edge of the clock. If the sequence of digital data words provided to the DAC represents samples of a sine wave, then the analog output is a piecewise constant function  110  that approximates a sine wave as shown. 
     A DAC may have a clock or “sampling clock” input, a digital data input word, and an analog output. The signal at the analog output may be updated once per clock cycle of the sampling clock. Any frequency at the analog output of the DAC may fall into a Nyquist zone, each of which spans a frequency range equal to one half of the sampling clock frequency, or “sampling frequency”. The first Nyquist zone is defined to be the frequency range from 0 (DC) to one-half of the sampling frequency, the second Nyquist zone is defined to be the frequency range from one-half of the sampling frequency to the sampling frequency, the third Nyquist zone is defined to be the frequency range from the sampling frequency to three-halves of the sampling frequency; further Nyquist zones are defined in an analogous manner, with the n th  Nyquist zone extending from (n−1)*f s /2 to n*f s /2, where f s  is the sampling frequency (which may also be referred to as f ox ). The analog output signal may have a component, referred to as the fundamental, in the first Nyquist zone, and other tones at other frequencies. The image in the second Nyquist zone, for example, is referred to as the first image, the image in the third Nyquist zone is referred to as the second image; in general the image in the n th  Nyquist zone is referred to as the (n−1) th  image. The maximum available frequency for the fundamental tone may be the Nyquist frequency, i.e., one-half of the sampling frequency. For applications in which a subsequent circuit (receiving the analog output of the DAC) requires an input signal at a higher frequency than the Nyquist frequency, a suitable Nyquist image may be used, instead of the fundamental, by the subsequent circuit. 
     Referring to  FIG. 2 , if the digital data stream supplied to the DAC consists of samples of a sine wave, the analog signal may have a spectrum consisting of a fundamental tone, and a plurality of images, all of which fall on an envelope  210  that is a sinc function of the sampling frequency. 
     For applications in which a Nyquist image is to be used, DAC output with a different envelope, e.g., an envelope providing more power to the Nyquist image to be used, may be generated by operating the DAC in another mode of several modes referred to herein as return to zero (RTZ) mode, inverse non return to zero (INRZ) mode, and inverse return to zero (IRTZ) mode. 
     Referring again to  FIG. 1 , in the RTZ mode, the analog signal at the analog output of the DAC is a piecewise constant function  120  equal the input data signal (e.g., a sine function) during half of each clock cycle, and equal to zero during the remainder of every clock cycle, as illustrated for the waveform  120  labeled “RTZ”. The envelope function for this mode may be the RTZ envelope  220  of  FIG. 2 . In the INRZ mode, the analog signal at the analog output of the DAC is a piecewise constant function equal to the input data signal (e.g., the sine function) during half of each clock cycle, and equal to the opposite of the input data signal during the remainder of every clock cycle, as illustrated for the waveform  130  labeled “INRZ”. The envelope function for this mode may be the INRZ envelope  230  of  FIG. 2 . In IRTZ mode, the analog signal at the analog output of the DAC is a piecewise constant function  140  equal to the input data signal (e.g., the sine function) during a first portion (e.g., one quarter) of each clock cycle, equal to the opposite of the input data signal during a second portion (e.g., one quarter) of each clock cycle, and equal to zero during a third portion (e.g., one half) of each clock cycle. The envelope function for this mode may be the IRTZ envelope  240  of  FIG. 2 . In one embodiment the RTZ, INRZ and IRTZ modes may each be selected when an output frequency is to be produced in a respective region of operation  225 ,  235 ,  245  shown in  FIG. 2 . This region of operation is where the output signal has the highest power level in the frequency spectrum according to the transfer function described in  FIG. 2 . In the NRZ conventional DAC mode, the signal power peaking at DC with nulls at n*Fs in  210  is where the DAC typically operating in the first Nyquist zone. In the RTZ DAC mode, the signal power also peaks at DC but with nulls at 2n*Fs in  220  is where the DAC typically operating in first or second Nyquist zones in region  225 . In the INRZ DAC mode, the signal power peaking in the second Nyquist zone with nulls at m*Fs where m=0, 2, 4, . . . in  230  is where the DAC optimally operating in the second Nyquist zone in region  235 . In the IRTZ DAC mode, the signal power peaks in the third Nyquist zone with nulls at m*Fs where m=0, 4, 8, . . . in  240  is where the DAC optimally operating in the third and fourth Nyquist zones in region  245 . Note that changing the sampling clock or signal inversion duty cycle will slightly alter the shape of the corresponding transfer function but the general characteristic remains the same. 
     In one embodiment a DAC configured to operate in four modes, i.e., in an NRZ mode, an RTZ mode, an INRZ mode, and an IRTZ mode may be constructed from a plurality of DAC cells, each of which may be constructed according to the schematic of  FIG. 3 . In the circuit of  FIG. 3 , a first control stage referred to as the RTZ control stage  310  is configured to operate in either of two states. This stage includes two differential pairs, each including two transistors with a common connection referred to as the tail of the differential pair. For example, in  FIG. 3 , each differential pair consists of two n-channel metal-oxide semiconductor field effect transistors (MOSFETs), the sources of the MOSFETs being connected together to form the tail of the differential pair. The two differential pairs are current steering pairs with the inputs of each pair connected in opposite polarity controlled by the differential RTZ control signal. The transistor  312  on the left in the left differential pair and the transistor  314  on the right in the right differential pair are “output transistors”  312 ,  314  of the DAC cell, that are connected to respective current outputs  320 ,  322  of the DAC cell. The RTZ control stage  310  turns off both output transistors  312 ,  314  when the RTZ control input is high, and turns on both output transistors when the RTZ control input is low. When the output transistors are turned on, currents flowing at two current inputs  316 ,  318  are conducted by the output transistors  312 ,  314  to respective outputs  320 ,  322  of the DAC cell (which may be referred to, with reference to the circuit of  FIG. 3 , as the positive current output of the DAC cell and the negative current output of the DAC cell, respectively). The outputs  320 ,  322  of the DAC cell may represent a differential output, i.e., a current of a first magnitude flowing at the first output  320  may represent a positive output signal of a first magnitude, and the same current flowing instead at the second output  322  may represent a negative output signal of the same magnitude. When the output transistors  312 ,  314  are turned off, grounding transistors  313 ,  315  are turned on, and the currents flowing at the two current inputs  316 ,  318  are conducted by the output transistors  312 ,  314  to Vdd or AC ground. When the signal current is steered to AC ground instead of used at the signal output, the current is effectively forming the return to zero function. 
     A second stage of the DAC cell  300  referred to as the data stage  330  includes two cross-coupled differential pairs, each consisting of two n-channel MOSFETs. When the data signal is high, currents flowing at first and second current inputs  332 ,  334  of the data stage are conducted to respective first and second current outputs  316 ,  318  of the data stage (which are the current inputs  316 ,  318  of the RTZ control stage  310 ). When the data input signal is low, currents flowing at first and second current inputs  332 ,  334  of the data stage are conducted to respective second and first current outputs  316 ,  318  of the data stage. Thus, when the data input is low, the output of the DAC cell is inverted, relative to the output it would generate if the data input were high. The second data stage is constructed such that both the normal polarity of the signal current and the inverted polarity of the signal are available to be steered to the DAC output. 
     A second control stage of the DAC cell, referred to as an invert control stage  340 , directs a current from a current input  342  to the first current input  332  of the data stage if its control input is high, or to the second current input  334  of the data stage if its control input is low. This control stage selects the normal polarity of the signal current and the inverted polarity of the signal current to be steer to the DAC output. The current input  342  may be connected to a current source or “current reference”  350 , which sources or sinks a current of a substantially fixed magnitude. 
     Each of the RTZ control stage  310 , the data stage  330 , and the invert control stage  340  may have a differential input, which may be connected to two or more transistor gates in the respective stage as shown. Each such differential input may include two conductors referred to as a positive conductor and a negative conductor, or as carrying a positive signal and a negative signal. This terminology indicates the differential nature of the signals and does not imply, for example, that one conductor carries a positive voltage and the other carries a negative voltage. Each of the three stages as shown has one or more current inputs (at the bottom of the respective stage, as illustrated in the schematic diagram of  FIG. 3 ) and two current outputs (at the top of the respective stage, as illustrated in the schematic diagram of  FIG. 3 ). This terminology (of designating certain conductors as current inputs or outputs) does not indicate the direction of current flow, but rather the sequence in which the stages are connected together, between the current reference and the outputs  320 ,  322  of the DAC cell. 
     Multiple DAC cells may be combined as shown in  FIG. 4 . In embodiments in which all DAC cells switch substantially the same current, the output current to voltage network  410  may be a pair of termination resistors, or a R2R network. In the former case, the DAC cells may be configured to form a unary DAC (and driven by a parallel unary data signal), with, for example, at any point in time, the number of DAC cells with the positive output turned on being proportional to the analog signal at that point in time. In this case the positive outputs from all of the DAC cells may be connected together, so that their currents are summed, and the total current may flow through a first termination resistor, e.g., to Vdd or equivalent AC ground. The negative outputs from all of the DAC cells may be also be connected together, so that their currents are summed, and the total current may flow through a second termination resistor to AC ground (e.g., to ground or to Vdd). The negative and positive outputs of the DAC cells may also be connected to respective terminals of a differential amplifier, so that the difference between voltages generated across the two termination resistors generates the output voltage at the output of the differential amplifier. 
     In another embodiment the DAC cells may be configured to form a binary DAC and driven with a binary data signal. In such an embodiment the output current to voltage network  410  may include two R2R resistor networks. The positive outputs of the DAC cells may be connected to a first R2R resistor network, the voltage at the output of which is a weighted sum of the DAC cell output currents, with the weights being a geometric progression with a ratio of 2. The negative outputs of the DAC cells may similarly be connected to a second R2R resistor network. 
     In another embodiment in which DAC cells may be configured to form a binary DAC and be driven with a binary data signal, the current sources of the DAC cells may provide different amounts of current, the current magnitudes forming a geometric progression with a ratio of 2 (i.e., each current source providing half as much current or twice as much current as another current source). In this embodiment the current to voltage network  410  may be a pair of termination resistors connected between (i) the two respective common nodes at which the pairs of conductors of the differential DAC current outputs are connected together, and (ii) AC ground. 
     In some embodiments these configurations may be combined. For example, a 12-bit compound DAC may be formed by 32 DAC cells configured as a 5-bit unary DAC, combined with 7 DAC cells configured as a 7-bit binary DAC. The 7 bit binary DAC may provide the most significant 7 bits of the compound DAC or the least significant 7 bits of the compound DAC. 
     A control circuit  420  may supply a differential RTZ control signal, a differential data signal, and a differential invert control signal to each of the DAC cells. The control circuit may be configurable, e.g., by a system-level controller, to operate in any of the four modes (NRZ, RTZ, INRZ, or IRTZ), or in a variant of IRTZ with a duty cycle different from 50%, discussed in further detail below. 
       FIG. 5  shows control waveforms that may be used when the DAC is operating in NRZ mode. The RTZ control signal is constant, in the low state, and the invert control signal is constant in the high state.  FIG. 6  shows control waveforms that may be used when the DAC is operating in RTZ mode. The RTZ control signal alternates between low and high at the same rate as the clock, and the invert control signal is constant in the high state.  FIG. 7  shows control waveforms that may be used when the DAC is operating in INRZ mode. The RTZ control signal is constant, in the low state, and the invert control signal alternates between low and high at the same rate as the clock.  FIG. 8  shows control waveforms that may be used when the DAC is operating in IRTZ mode. The RTZ control signal alternates between low and high at the same rate as the clock, and the invert control signal alternates between low and high at twice the rate of the clock. 
     Referring to  FIG. 9 , in a modified-duty-cycle IRTZ mode, the RTZ control input is low for a part (e.g., the first ⅔) of the each clock cycle and high for the remainder of each clock cycle, and the invert control input is high for a part (e.g., the first ⅓) of the each clock cycle and low for the remainder of each clock cycle. The analog output waveform then is a piecewise constant function equal to the input data signal (e.g., the sine function) during a first portion of each clock cycle, equal to the opposite of the input data signal during a second portion of each clock cycle, and equal to zero during a third portion of each clock cycle. In the example of  FIG. 9 , each of the first portion, the second portion, and the third portion has a duration of one third of a clock cycle; in other embodiments the relative durations of the three portions may be adjusted arbitrarily by adjusting the waveforms at the RTZ control input and the invert control input accordingly. The use of this mode may improve the spurious-free dynamic range, or “spur-free dynamic range” of the DAC, by providing an increased settling time after each transition. 
     Although the circuit of  FIG. 3  employs n-channel MOSFETs, in other embodiments other transistors, e.g., p-channel MOSFETs or bipolar junction transistors (BJTs) may be used instead of, or in combination with n-channel MOSFETs in any of the processing technologies such as Silicon, Silicon Germanium, GaAs, GaN, InP, etc. For example,  FIG. 10  shows a DAC cell analogous to the cell illustrated in  FIG. 3 , constructed using NPN bipolar junction transistors. 
     Although limited embodiments of a wideband multi-mode current switch for digital to analog converter have been specifically described and illustrated herein, many modifications and variations will be apparent to those skilled in the art. Accordingly, it is to be understood that a wideband multi-mode current switch for digital to analog converter employed according to principles of this invention may be embodied other than as specifically described herein. The invention is also defined in the following claims, and equivalents thereof.