Abstract:
An apparatus for measuring emission time delay during irradiation of targeted samples by utilizing digital signal processing to determine the emission phase shift caused by the sample is disclosed. The apparatus includes a source of electromagnetic radiation adapted to irradiate a target sample. A mechanism generates first and second digital input signals of known frequencies with a known phase relationship, and a device then converts the first and second digital input signals to analog sinusoidal signals. An element is provided to direct the first input signal to the electromagnetic radiation source to modulate the source by the frequency thereof to irradiate the target sample and generate a target sample emission. A device detects the target sample emission and produces a corresponding first output signal having a phase shift relative to the phase of the first input signal, the phase shift being caused by the irradiation time delay in the sample. A member produces a known phase shift in the second input signal to create a second output signal. A mechanism is then provided for converting each of the first and second analog output signals to digital signals. A mixer receives the first and second digital output signals and compares the signal phase relationship therebetween to produce a signal indicative of the change in phase relationship between the first and second output signals caused by the target sample emission. Finally, a feedback arrangement alters the phase of the second input signal based on the mixer signal to ultimately place the first and second output signals in quadrature. Mechanisms for enhancing this phase comparison and adjustment technique are also disclosed.

Description:
CROSS REFERENCE TO RELATED APPLICATION 
   This patent application is a division of presently U.S. patent application Ser. No. 09/205,755, filed Dec. 4, 1998 now U.S. Pat. No. 6,157,037, the contents of which are specifically incorporated herein by reference. 

   CONTRACTUAL ORIGIN OF THE INVENTION 
   This invention was made with U.S. Government support under contract NAS9-97080 awarded by NASA and contract F33615-97-0729 awarded by the Department of the Air Force. The Government has certain rights in this invention. 

   BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   This invention relates generally to sensing instruments and methods for measuring the concentration of an analyte in a medium and, more particularly, to a device and method for measuring such concentrations by measuring the emission time delay during irradiation of a targeted sample surrounded by the analyte. Specifically, the present invention relates to a device and method for measuring exponential time constants, phase shifts, time delays and parameters derivable therefrom caused by irradiation of a targeted sample utilizing digital signal processing and especially luminescence quenching systems, phase shifts through networks, and time delays of photon migration through media. 
   2. Description of the Prior Art 
   Dynamic phase modulation, quenched luminescence sensors are well known. Instruments of this type have been, for example, developed or proposed for use in hospitals to monitor the concentration of gases such as oxygen, ionized hydrogen and carbon dioxide within the blood of patients. The particular substance of interest, for example oxygen, is known as the analyte. 
   As is known in the art, luminescence materials absorb energy and are driven from their ground state energy level to an excited state energy level in response to the application of energy from an electromagnetic radiation source such as light. These materials are unstable in their excited states, and they luminesce or give off excess energy as they return to their ground state. For example, the short wavelength ultraviolet light of black light stimulates dyes in a colored fabric to emit longer wavelengths, such as blue, green or red, and thus fluoresce. For the purposes of the present disclosure, the term “luminescence” as used herein is a general term which describes both fluorescence and phosphorescence, for all three terms are frequently used interchangeably in the art. The distinction and overlap of the terms is obvious to one skilled in the art. 
   In the presence of certain chemicals, many fluorescent materials are said to be quenched, i.e. the time constant of the fluorescence emission is altered by the effects of the surrounding chemicals. The degree of quenching of the fluorescence in turn can be related to the concentration of the quencher, which for example may be a chemical dissolved in water or mixed in air, such as oxygen in the blood of patients as explained above. There is a substantial amount of literature that describes fluorescent molecules that are selectively quenched by oxygen, carbon dioxide, glucose, pH, NH 3 , metal ions, temperature and other environmentally and medically important analytes. These analytes are relevant to applications such as monitoring drinking water quality, industrial process control, monitoring of human respiratory function, human blood analysis for critical care patients, and the like. 
   One of the obstacles to the commercialization of fluorescence sensing devices has been a lack of inexpensive yet accurate instrumentation for the measurement of changes in the fluorescent time constant. For example, U.S. Pat. Nos. 4,845,368, 5,257,202, 5,495,850, 5,515,864 and 5,626,134 all disclose devices for measuring analyte concentration levels based on fluorescence. However, these particular devices are generally expensive and complicated. 
   The fluorescence lifetime or time constant, τ is the amount of time it takes the fluorescence emission to decrease by a factor of 1/e or about 63% after termination of irradiation as disclosed in U.S. Pat. No. 4,716,363 by Dukes et al, in column 1, lines 37-41. This is common knowledge and is available in the literature reference, i.e. “Topics in Fluorescence Vol. 2—Principles”, ed. Joseph Lakowicz. If light modulated sinusoidally at a frequency, ƒ, is thus applied to the fluorescence sensor, the output is a sinusoidal emission of identical frequency, but having a phase shift and reduced amplitude with respect to the excitation signal. The equation governing the relationship between modulation frequency, ƒ, phase shift, θ, and the fluorescent time constant, τ, is as follows: 
             τ   =           tan   ⁢           ⁢   θ       2   ⁢   π   ⁢           ⁢   f       ⁢           ⁢   or   ⁢           ⁢   θ     =     arctan   ⁡     (     2   ⁢   π   ⁢           ⁢   f   ⁢           ⁢   τ     )                 (     Equation   ⁢           ⁢   1     )             
 
or
 
θ=arctan(2πƒτ)   (Equation 1) 
 
   Thus, if we know the excitation modulation frequency and can measure the phase shift of the emission signal relative to the excitation signal, we can determine the fluorescence constant, τ, using the above Equation 1. In a fluorescence-based sensor, the fluorescence time constant is measured since this fluorescence time constant is altered by the presence of certain chemical species. Consequently, the concentration of chemical species can be determined by measuring the fluorescence time constant by measuring the phase shift associated therewith. 
   According to Equation 1, in order to measure the fluorescence time constant, one must know the excitation modulation frequency, ƒ, and the phase shift of the light through the fluorescence system. With these quantities, the fluorescence time constant can be calculated and then related to analyte concentration. There are several different known techniques for determining the excitation frequency and phase shift of a system with an unknown time constant. One manner of determining this is by exciting the sample with a fixed frequency signal and then measuring the phase shift that results, that is the sample excitation modulation frequency is maintained constant while the signal phase, which varies with analyte concentration, is measured. U.S. Pat. Nos. 5,317,162, 5,462,879, 5,485,530 and 5,504,337 all disclose such fixed frequency, variable phase techniques and devices. Of particular interest is an article by Venkatesh Vadde and Vivek Srinivas entitled, “A closed loop scheme for phase-sensitive fluorometry”, the American Institute of Physics, Rev. Sci. Instrum., Vol. 66, No. 7, July 1995, p. 3750. 
   Another principal way of conducting the above measurements is by exciting the sample with a modulation frequency that maintains a constant phase relationship between the excitation signal and the emission signal, that is the excitation frequency is varied in order to maintain a particular phase relationship. Such devices and techniques are known as phase-modulation, fluorescence-based sensing devices and are clearly illustrated in U.S. Pat. Nos. 4,840,485, 5,196,709, and 5,212,386, and in an article by Brett A. Feddersen, et al. entitled, “Digital parallel acquisition in frequency domain fluorimetry”, American Institute of Physics, Rev. Sci. Instrum., Vol. 60, No. 9, September 1989, p. 2929. Of particular interest in U.S. Pat. No. 4,716,363 by Dukes et al., which describes a feedback system that provides the modulation frequency required to give a constant phase shift of about 45°. The resulting frequency is then used to determine the analyte concentration which is a function of excited state lifetime. 
   U.S. Pat. No. 5,818,582 teaches the use of a DSP for fluorescence lifetime measurements, though not using quadrature signal comparison for determination of fluorescent sample phase shifts. 
   Despite the availability of the above-discussed techniques and sensing devices, there is a continuing need for improved fluorescence-based sensing instruments. In particular, there is a need for such devices which are useful for a broad range of applications involving exponential decay and time delay measurements, which are made from inexpensive components, and which present measurements in real time without the need for off-line signal processing as is the case of the patents to Federson, Gratton and others. A major detriment to many of the devices presently available is that they are very expensive to acquire and maintain. Moreover, analog systems of the present art are subject to drift and therefore unnecessary errors. Such systems should be, to the contrary, inexpensive, convenient to use and provide adequate sensitivity over an extended and continuous measurement range. The system of the Dukes patent emphasizes optimal sensitivity over a wide measurement range, but in so doing, requires very complex and expensive system components. To the contrary, optimal sensitivity can be sacrificed for sub-optimal, adequate sensitivity in order to achieve inexpensive, less complicated measurement techniques. In addition, the measurement approach of such devices should be susceptible to convenient and precise readout. 
   SUMMARY OF THE INVENTION 
   Accordingly, it is one object of the present invention to provide an apparatus and method for measuring emission time delay during irradiation of targeted samples. 
   It is another object of the present invention to provide sensing instruments which are applicable to a broad range time delay, phase shift and exponential decay measurements involving luminescent materials and various scattering media. 
   Yet another object of the present invention is to provide fluorescence-based sensing instruments which are made from inexpensive components. 
   Still another object of the present invention is to provide an apparatus and method for measuring emission time delay during irradiation of targeted samples utilizing digital signal processing to determine the emission phase shift caused by the sample. 
   A further object of the present invention is to provide an apparatus and method for measuring luminescence-quenching systems, specifically oxygen sensitive systems. 
   To achieve the foregoing and other objects and in accordance with the purpose of the present invention, as embodies and broadly described herein, an apparatus is disclosed for measuring emission time delay during irradiation of targeted samples by utilizing digital signal processing to determine the emission phase shift caused by the sample. The apparatus includes a source of electromagnetic radiation adapted to irradiate a target sample. A mechanism generates first and second digital input signals of known frequencies with a known variable phase relationship, and a device then converts the first and second digital input signals to analog sinusoidal signals. An element is provided to direct the first input signal to the electromagnetic radiation source to modulate the source by the frequency thereof to irradiate the target sample and generate a target sample emission. A device detects the target sample emission and produces a corresponding first output signal having a phase shift relative to the phase of the first input signal, the phase shift being caused by the emission time delay in the sample. A member produces a known phase shift in the second input signal to create a second output signal. A mechanism is then provided for converting each of the first and second analog output signals to digital signals. A mixer receives the first and second digital output signals and compares the signal phase relationship therebetween to produce a signal indicative of the change in phase relationship between the first and second output signals caused by the target sample emission. Finally, a feedback arrangement alters the phase of the second input signal based on the mixer signal to ultimately place the first and second output signals in quadrature. Mechanisms for enhancing this phase comparison and adjustment technique are also disclosed. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The accompanying drawings which are incorporated in and form a part of the specification illustrate preferred embodiments of the present invention and, together with a description, serve to explain the principles of the invention. In the drawings: 
       FIG. 1  is a schematic illustrating an embodiment of the present invention utilizing a direct phase adjustment, constant frequency technique with digital signal processing for measuring emission phase shift to determine time delay through an irradiated sample. 
       FIG. 2  is a schematic illustrating another embodiment of the present invention utilizing variable-frequency and variable-phase techniques for measuring emission phase shift to determine time delay through an irradiated sample. 
       FIG. 3  is a schematic illustrating yet another embodiment of the present invention similar to that of  FIG. 2  but incorporating signal down conversion steps. 
       FIG. 4  is a schematic illustrating yet another embodiment of the present invention similar to that of  FIG. 3  but incorporating dual quadrature signal down conversion steps. 
       FIG. 5  is a schematic illustrating yet another embodiment of the present invention similar to that of  FIG. 4  but eliminating the means for the downconverting of high frequency signals to lower frequencies for quadrature phase detection. 
       FIG. 6  is a schematic illustrating yet another embodiment of the present invention using a single analog timing element and a DSP for real time determination of phase and lifetime. 
   

   DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
   Referring initially to  FIG. 1 , a closed-loop lifetime measurement device  10  is illustrated which incorporates a Digital Signal Processor (DSP)  12 . Venkatesh and Srinivas, in the prior art references discussed above, disclose a closed-loop fluorescent-decay time measurement system that allows a phase demodulator to operate in an optimal null condition. This is accomplished using two analog timing elements including a crystal-controlled frequency generator and a second analog timing element to vary the phase of a measurement reference signal so that signals to the demodulator are always in quadrature. An important aspect of this disclosed technique is that the reference channel is phase advanced in order to maintain quadrature at the phase demodulator. 
   Within the DSP of  FIG. 1 , all the components referred to below are digital and defined by software loaded into the DSP chip. In the DSP  12 , a constant frequency, dual-output, variable-phase waveform generator  14  is provided and adapted to generate a first experimental digital signal  16  and a second reference digital signal  26 . The digital signals  16 ,  26  are preferably identical in frequency. The experimental signal  16  is directed through a digital-to-analog converter  20  where it is converted into an analog signal which drives an electromagnetic radiation-emitting device  22 . The dual-output, variable-phase waveform generator  14  causes the reference signal  26  to be phase advanced a known number of degrees (N°) relative to experimental signal  16 . Reference signal  26  is then directed through a second digital-to-analog converter  28 . 
   In preferred form, the device  22  is a light emitting diode (LED). The device  22  is activated by the analog form of the signal  16  and generates a light emission  30 . The emission  30  is directed to a target sample  32  which includes therein material which will emit energy  34  as a result of being impinged by the light  30 . In one preferred form of the invention, the target sample  32  is a fluorescent material designed to generate a fluorescence emission  34  upon contact with the LED emission  30 . However, it should be understood that the target sample  32  may include any appropriate emission-delay generating system as discussed above. 
   The emissions  34  are detected by a device  36 , which in preferred form is a photodiode. The detection device  36  then generates an output signal  38  which has a frequency identical to the experimental signal  16  but is phase retarded as a result of the time delay imposed by the target sample  32 . Thus, the output signal  38  is now phase-shifted an unknown amount relative to the experimental signal  16  and the reference signal  26 . In preferred form, the output signal  38  is directed through a pre-amplifier  40 . A pair of anti-aliasing filters  42 ,  44  are provided, and the output signal  38  passes through the filter  42  while the reference signal  26  passes through the filter  44 , thereby becoming output reference signal  46 . Thus, both the experimental signal  16  and reference signal  26  are effectively treated substantially identically outside the DSP  12  except for the phase shift resulting from the target sample  32 . This is due to the fact that while the LED  22 , the photodiode  36  and the pre-amplifier  40  all add phase shifts to the experimental signal  16  that do not exist in the reference signal  26  chain, these shifts, though significant, are calibrated out. In the preferred embodiment, the phase shift caused by LED  22 , the photodiode  36  and the pre-amplifier  40  is negligible compared to the phase shift caused by anti-aliasing filters  42  and  44 . The system  10  uses duplicate anti-aliasing filters  42 ,  44  to eliminate major phase imbalances that would otherwise exist in the two channels. Both signals  38  and  46  are passed through respective analog-to-digital converters  48 ,  50  to create counterpart digital output signals within the DSP  12 . 
   The output reference signal  46  is then mixed with the output experimental signal  38  at the signal mixing device  52 , which in this particular embodiment is preferably a phase demodulator. As was previously stated, the original reference signal  26  is phase-shifted by the digital waveform generator  14  a specific number of degrees so that the experimental input signal  16  and the reference input signal  26  are out-of-phase by a known, predetermined amount. However, due to the phase shift imposed by the target sample  32 , the phase differences between the output signals  38  and  46  at the mixing member  52  are unknown. Therefore and in preferred form, the phase demodulator  52  indicates when the reference signal  46  (A) relative to the experimental signal  38  (B) are in quadrature, or 90° apart. In other words, 90°−B=A if the signals are in quadrature. If the signals  38  and  46  are in fact in quadrature, then no changes are made to the relative phase offset between input signals  16  and  26 . However, due to the phase shift caused by the target sample  32 , the signals  38  and  46  are not initially in quadrature at  52 . 
   As a result, the phase demodulator  52  generates a signal  54  comprised of both AC and DC components, the DC component represents the phase difference between the signals  38  and  46  relative to 90°. This signal  54  is preferably passed through a low pass filter  56  to remove the AC component creating a DC error signal  58 . The sign and magnitude of the DC error signal  58  indicates the relative phase difference between input signals  38  and  46  and is preferably zero when the input signals  38  and  46  are in quadrature. Based on the error signal  58 , the digital waveform generator  14  continuously modifies the phase advance (N°) of the reference signal  26 . In this manner, the device  10  continues to change the relative phase of the signals  16  and  26  until the phases of the output experimental signal  38  and the output reference signal  46  are in quadrature at the phase demodulator  52 , at which point error signal  58  is substantially zero. At this stage, the phase shift through the sample  32  is 90°−N°, with N° being the phase advance of the signal  26  relative to signal  16 . This phase shift quantity is then utilized with the known frequency of the experimental signal  16  and Equation 1 to calculate the lifetime of the target sample  32 , which in turn will provide the desired information about the analyte surrounding the target sample  32  as discussed above. 
   One of the important aspects of the embodiment illustrated in  FIG. 1  is that this embodiment utilizes parallel analog paths for both the reference and experimental signals in combination with digital processing. These parallel paths are used for two principal reasons. The first is that the digital-to-analog converters  20 ,  28  as well as the analog-to-digital converters  48 ,  50  introduce time delays into the signals passing through them. Any difference in time delay between the two paths will result in an undesired phase-offset between them. However, if both of the experimental and reference signals  16 ,  26  pass through matched identical converters  20 ,  28  and  48 ,  50 , the reconstruction and digitization will not result in a relative time delay of one signal with respect to the other. 
   The second reason is that the anti-aliasing filter  42  introduces a significant amount of phase lag into the experimental analog signal  38 , i.e. about 26 degrees at 20 kHz. By passing the reference analog signal  26  through an identical anti-aliasing filter  44 , the phase lag as a result of the anti-aliasing filter is canceled. Additionally, any drift in phase cause by the anti-aliasing filter  42  in the experimental signal path  38  will tend to be canceled by similar drift in the anti-aliasing filter in the reference signal path  26 . The symmetrical treatment of the experimental  38  and reference  46  signals means that the phase difference between them is due only to the phase delay resulting from the target sample  32  as well as the known phase advance created by the digital phase shifter  24 . While the LED driver circuit, the LED, the photodiode and the preamplifier all contribute small phase shifts in the experimental signal  16  that are not cancelled by similar components in the reference signal  26  path, these phase shifts are cancelled in other ways as indicated above. In addition, since all operations within the DSP  12  are digital calculations, they are free from any drift or non-linearity whatsoever. 
   In the above embodiment of  FIG. 1 , the DSP  12  implementation of the device  10  utilizes only one analog timing element for the generation of the reference and experimental signals  26 ,  16 , for the basic phase shifting of the reference signal  26 , and for the phase demodulation of signals  46  and  38  at the demodulator  52 . The time base for the DSP frequency generation and the phase shifting is preferably derived from a single external crystal oscillator. Moreover, the phase shifting of the reference signal  26  is preferably accomplished by the addition of two 32-bit numbers which does not introduce a phase jitter as is true of pure analog systems or of systems using more than one analog timing element for signal generation, comparison and phase shifting. As a result of the lack of phase jitter or instability (drift) between the two signals  16 ,  26 , extremely small phase changes, e.g. 0.001 degrees, caused by the target sample  32  are detectable by the device  10 . Moreover, the advantage of using identical anti-aliasing filters  42 ,  44  is that any changes in the filter properties resulting from changing temperatures are reflected in both the experimental and reference signal paths and are therefore canceled. This substantially reduces electronic phase drift as compared to prior art devices. 
   As was discussed above, existing devices also utilize changes in frequency, rather than phase, to measure fluorescence emissions or the like. In particular, U.S. Pat. No. 4,716,363 by Dukes et al. describes a fluorescence lifetime measurement system that operates in this manner. In particular, the excitation frequency of Dukes is varied such that a constant predetermined phase shift is obtained through the fluorescence experiment. The predetermined phase shift is selected to achieve optimal sensitivity to changes in lifetime. Since the frequency is inversely proportional to the lifetime as illustrated above in Equation 1, the frequency can be directly related to the quencher or analyte concentration of the target sample  32  thereby circumventing the need to calculate lifetime or phase. 
   While this particular system of Dukes operates fine in certain instances, there are significant drawbacks. Without going into a detailed discussion of this reference, the Dukes&#39; system operates such that the oscillator frequency is adjusted to maintain a fixed and optimum phase-shift through the fluorescence experiment. However, in sensing applications where the change in the lifetime of the fluorescent experiment is large, the frequency must change over an equally large range. Thus, if the lifetime changes by a factor of 100, then the oscillator must change frequency by a factor of 100. The use of smaller or larger phase-offsets will shaft the maximum and minimum frequencies up and down but will not compress the required range. There are many situations where generating frequencies over such a wide range is impractical because it is prohibitively complex or expensive. As a result of this problem, the present invention provides the additional embodiments of the invention as illustrated in  FIGS. 2-4 . 
   Referring now in particular to  FIG. 2 , it should be understood that like components throughout all of the Figures and embodiments of the present invention will have like numerals and indicators. In this particular embodiment of  FIG. 2 , the device  10  includes a DSP  60  preferably includes a dual-output, variable-phase waveform generator  62 . Within the waveform generator  62  is a frequency and phase calculator  64  which determines the appropriate frequency and phase relationship of the two signals  16  and  26  output by generator  62 . The frequency generator  62  generates an initial experimental signal  16  with a known phase and an initial reference signal  26  which is phase advanced relative to the experimental signal  16  by the calculator  64 . 
   As with the prior embodiment of  FIG. 1 , the signals  16 ,  26  pass through their respective digital-to-analog converters  20 ,  28 , and the experimental signal  16  activates an electromagnetic radiation emitting device  22  such as an LED. The emissions  30  impinge on the target sample  32  which in turn generates emissions  34  detected by the detection member  36  such as a photodiode, all being similar to the prior embodiment illustrated in FIG.  1 . The experimental output signal  38  passes through a pre-amplifier  40 , and both the experimental output signal  38  passes through a pre-amplifier  40 , and both the experimental output signal  38  and the reference output signal  26  pass through respective anti-aliasing in filters  42 ,  44  and analog-to-digital converters  48 ,  50 . As in the prior embodiment, the output signals  38 ,  46 , in digital form, are then combined at the mixer  52 . If the signals  38 ,  46  are not in quadrature, the DC component of the signal  54  is a number other than zero. 
   The AC component of signal  54  is removed by the low pass filter  56 . The output of the filter  56  is a DC error signal  58 , the sign and magnitude of which indicates the relative phase difference between signals  46  and  38 . As explained previously, the DC error signal  58  is preferably zero when signals  38  and  46  are in quadrature. The DC error signal  58  output from the filter  56  is then directed back to the frequency calculator  64  within the waveform generator  62  to simultaneously control both the frequency and the phase of the output signals  16  and  26  as described below. 
   In this embodiment of  FIG. 2 , the feedback error signal  58  causes the waveform generator  62  to simultaneously change both the phase advance of the signal  26  relative to the signal  16  as well as the modulation frequency of both signals  16  and  26 . The phase and modulation frequency are changed simultaneously until the DC error signal  58  indicates that input signals  46  and  38  are in quadrature, that is when the error signal  58  is substantially zero. The phase and frequencies of the waveforms determined by the calculator  64  are indicated by binary numbers stored in the DSP  60 . Although the waveform generator  62  and the contained frequency and phase calculator  64  are digital, the high digital resolution affords effectively continuous changes in frequency and phase offset. Thus, when the DC error signal  58  is substantially zero, the time constant of the luminescence system can be calculated using Equation 1. The phase delay through the sample  32  is simply 90°−N° with N° being the phase advance of the signal  26 , and the frequency is known from the digital number generated by the calculator  64 . 
   The simultaneous and continuous variation of phase and frequency in a feedback loop acts to compress the phase and frequency ranges that are required for a particular luminescence lifetime range. Compared to the prior art techniques of Dukes and Venkatesh as disclosed above, this  FIG. 2  embodiment of the invention uses less expensive, more convenient components that have narrower operating ranges. Moreover, while this embodiment of  FIG. 2  of the present invention employs continual and simultaneous changes in phase offset and frequency of the output signals  16  and  26 , thereby sacrificing optimum lifetime measurement sensitivity, a heretofore unanticipated result is the benefit of frequency and phase range compression for luminescence sensors. 
   This compression of the frequency and phase range over a wide lifetime range is accomplished by using the continuously variable phase offset and continuously variable frequency provided by the waveform generator  62 . Since the phase offset of the dual-output, variable-phase oscillator  62  changes as the frequency changes, then a much smaller frequency range is needed. For example, if the phase-offset of the variable-phase oscillator  62  were to change by 0.0038 degrees per Hz, then the entire range between 1 μsec and 100 μsec can be covered with a frequency range of 19,900 Hz-3,900 Hz. This is a frequency range of only 5:1 as compared to a range of 100:1 required by prior art devices and techniques using variable frequency and a predetermined, fixed phase offset. Thus, in this particular embodiment illustrated in  FIG. 2 , both the phase and frequency may vary and is known as “phase compression”, for the use of a continuously variable phase-offset compresses the frequency and phase ranges. This extends the measurable lifetime range for systems using inexpensive, limited range components including but not limited to oscillators, waveform generators, amplifiers, and analog-to-digital and digital-to-analog converters. 
   In the phase compression system  10  of  FIG. 2 , the frequency and phase output of the multiple phase oscillator  62  is determined by a DC error signal  58  derived from the mixer  52 . The error signal  58  from the mixer  52  and low pass filter  56  controls the output frequency and phase of the oscillator  62  so that the two signals  38 ,  46  input to the mixer  52  are eventually in quadrature. This error signal  58  passes to the frequency and phase calculator  64  which determines how the frequency and phase should change based on the error signal  58 . The outputs of the calculator  64  are binary numbers representing frequency and phase, and these numbers are used by the waveform generator  62  to generate a digital representation of two sine waves,  16  and  26 , at the frequency and phase offset specified by the calculator  64 . 
   While the waveform generator  62  creates a reference signal  26  that is advanced with respect to the experimental signal  16  as with the embodiment of  FIG. 1 , the difference in this embodiment of  FIG. 2  is that the frequency of the signals  16  and  26  change continuously and simultaneously with changes in the phase offset. In one preferred embodiment, the calculator  64  changes the frequency and phase according to the following relationship:
 
 N°=F·CF+N   base    Equation (2) 
 
where N° is the phase offset between signals  16  and  26 , F is the frequency of signals  16  and  26 , CF is the compression factor, and N base  is the base phase offset.
 
   For example with CF=0.0038 deg/Hz, N base =6.18 deg and a luminescence sensor lifetime of 101.6 μsec, the calculator  64  adjusts the system frequency to 4000 Hz, and the phase offset to 21°. These are the conditions where the error signal  58  is substantially zero. With equivalent CF and N base  parameters, and a luminescent sensor lifetime of 1.2 μsec, the calculator adjusts the system frequency to 19,900 Hz, and the phase offset to 82°. When the signals  38  and  46  are in quadrature, the lifetime of the sample  32  can be calculated using the known phase and frequency. Since the action of the calculator  64  compresses the frequency and phase range, inexpensive components with limited range can be used in the present invention in place of expensive and complex component. 
   In the above example, the frequency and phase range compression causes the shortest lifetime to require a frequency of under 20,000 Hz. Currently, it is a distinct advantage to use DSP compatible analog-to-digital and digital-to-analog converters that have a maximum frequency range of 20,000 Hz. This is due to the fact that such limited frequency range components are mass produced for consumer audio applications and thus are inexpensive and simpler to use as compared to wider frequency range components produced for more limited markets, e.g. scientific instrumentation. 
   A third embodiment of the present invention incorporates the concept of down conversion by mixing the reference and experimental signals with another third signal of different phase and frequency, i.e. down converting, to a fixed or variable low frequency while preserving relative phase information. In a simple lifetime measurement system embodiment as illustrated in FIG.  1  and  FIG. 2 , the frequency at which the exciting light  30  is modulated and the frequency at which phase measurement takes place in the DSP  12  are essentially identical. As higher modulation frequencies are demanded by the measurement of shorter fluorescent or luminescent lifetimes, a point is reached where the necessary program steps for phase comparison and correction cannot be executed between samples. This particular problem is overcome by the embodiments illustrated in  FIGS. 3-4 . In these embodiments, the high-frequency experimental and reference signals are each linearly multiplied by a local oscillator frequency in a mixer. The resulting waveform or signal is then filtered and presented to the analog-to-digital converter. 
   Referring now with particularity to  FIG. 3 , the device  10  includes a DSP  80  having a dual-output, variable-phase waveform generator  62 . As in the prior embodiment, a frequency and phase calculator  64  determines the appropriate frequency and phase relationship of the two signals  16  and  26  output by generator  62 . The experimental signal  16  and the reference signal  26  pass through their respective converters  20 ,  28 . The signal  16  activates an LED  22  which generates an emission  30  to impinge target sample  32  to create an emission  34  which is detected by the photodiode  36 . The output signal  38  passes though the pre-amplifier  40 . 
   In this particular embodiment, a second frequency generator  82  is disposed within the DSP  80  and generates a signal  84  having a frequency different from the frequencies of the output signals  38 ,  46 . The signal  84  passes through a digital-to-analog converter  86  and is then mixed with the output reference signal  46  at a mixer  88  as well as with the output experimental signal  38  at yet another mixer  90 . When the signal  84  mixes with each of the signals  38 ,  46 , a modified output reference signal  94  and a modified output experimental signal  92  are created, respectively. Each of the signals  92  and  94  passes through their respective anti-aliasing filters  44  and  42 , and analog-to-digital converters  50 ,  48  and are then demodulated at the digital mixer  52 . 
   When the signal  84  is mixed with the reference signal  46 , both the sum and the difference frequencies are incorporated into the modified output signal  92 . Likewise, both the sum and difference frequencies of the signal  84  and the signal  38  are reflected in the modified output signal  94 . The anti-aliasing filters  42  and  44  preferably remove the sum frequency of the signals  84  and  46  and the sum frequency of the signals  84  and  38 , respectively, so that only the difference frequency of the signals  84  and  46  and difference frequency of the signals  84  in  38  are mixed and compared at the mixer device  52 . At the demodulator  52 , the phases of the signals  92  and  94  are compared, and a feedback signal  54  is generated by the mixer  52 . This feedback signal  54  passes through the low pass filter  56  and is then returned to the frequency generator  82 . As in the prior embodiment, the error signal  58  indicates the sign and magnitude of the phase difference between signals  92  and  94 , and is preferably zero when these signals are in quadrature. The calculator  64  simultaneously changes the phase and frequency of output signal  16  and  26 , as in the prior embodiment of  FIG. 2 , such that a condition of quadrature is maintained at the mixer  52 . 
   The difference frequencies of the modified output signals  92  and  94  are held constant by action of the error signal  58  on the second frequency generator  82 . The second frequency generator  82  tracks the signal frequency output of the frequency generator  62  by always maintaining a signal frequency output that is different, i.e. higher or lower, by a constant value, for example 10 kHz. Constant frequency inputs to the demodulator  52  are preferred. One can anticipate a scheme, however, which sends variable frequency inputs to the demodulator  52  though there is generally no benefit to such an implementation. 
   In evaluating this embodiment of  FIG. 3 , when a sinusoid of one frequency linearly multiplies a sinusoid of another frequency, the resulting waveform or signal consists of a linear combination of a pair of sinusoids whose individual frequencies are the sum and difference of the two original frequencies. In a practical lifetime measurement circuit as in the embodiment of  FIG. 3 , the sum frequency is rejected by a filter, and the difference frequency which may be quite low is passed to the analog-to-digital converters for further processing within the DSP  80  as described above. The difference frequency can have any convenient value, and it is determined only by the relationship between the signals  38  and  46  frequency and the signal  84  frequency. The phase relationship between the high frequency reference signal  46  and the experimental signal  38  are maintained through the down conversion process. 
   Another embodiment of  FIG. 3  uses external digital waveform generators in place of the component generators  62  and  82  and the D/A converters  20 ,  28  and  86 . This particular embodiment would be used when the frequencies of the signals  16  and  26  are too high to be generated internally within digital signal processor  80 . In this case the external generators would preferably consist of single chip waveform generators which would be controlled by the digital signal processor  80  and derive their clock frequency from the same analog oscillator as digital signal processor  80 . 
   Referring now to  FIG. 4 , this embodiment imposes an additional requirement and capability on the down conversion process as compared to that of  FIG. 3  explained above. As a part of measuring the phase difference between the experimental and reference signal sinusoids, the DSP of this  FIG. 4  executes a program that implements an additional numerical direct digital synthesis frequency generator, the numerical direct digital synthesis generator being used in all the prior embodiments as the devices  14  and  64 . Referring to  FIG. 4 , the down conversion arrangement of  FIG. 3  remains substantially the same. However, the experimental signal  16  and the reference signal  26  are generated by one single-output frequency generator  96  and are generated at the same phase and frequency. 
   In one embodiment as illustrated in  FIG. 4 , the frequency generators  96  and  82  are specialized DSP components that are external to the main digital signal processor  98  and contain digital-to-analog converters  142  and  140 , respectively. In another embodiment, the frequency generators  96  and  82  may be internal to the main DSP  98 , as shown for example in  FIGS. 1 and 2 . While it is actually preferred that the frequency generator  96  is implemented internally within the DSP  98 , one would then preferably then use the downconversion single quad technique of FIG.  2 . When the signals  92  and  94  in this embodiment of  FIG. 4  pass through the analog-to-digital converters  50  and  48 , they are not mixed directly together as with the prior embodiments. Instead, an additional dual output, multiphase digital synthesis frequency generator  100  is provided within the DSP  98 . 
   In preferred form, the dual-output, multi-phase digital synthesis frequency generator  100  includes a frequency and phase calculator  102  that generates a first internal signal  104  and a second internal signal  106 , each of which has a frequency which matches exactly the frequency of the signals  92  and  94 , which is the difference frequency between the frequency generated by the generator  96  and the frequency generated by the generator  82 . The first internal signal  104  is generated such that it has a phase relative to input signal  92  of 90°. This is accomplished by mixing the signals  92  and  104  at an internal mixer  108 . The output of the mixer  108  is directed to a low pass filter  150  which outputs an error signal  152 , the sign and magnitude of which indicates the relative phase difference between signals  104  and  92 . The error signal  152 , preferably zero when the signals  92  and  104  are in quadrature, is directed to the frequency and phase calculator  102 . The calculator  102  then adjusts the phase of the signal  104  until the error signal  152  indicates that signals  92  and  104  are in quadrature. 
   As the same time, the signal  94  is directed toward another internal mixer  110 , and the frequency generator  100  generates the second internal signal  106  of preferably identical frequency with the signals  92 ,  94  and  104 , and with a phase that is advanced a known amount with respect to the phase of the signal  104 . The signal  106  is mixed with the signal  94  at the internal mixer  110 , the output of which is directed through a low pass filter  56  to create another internal error signal  114  which is directed to the frequency and phase calculator  102 . Based on the sign and magnitude of the error signal  114 , the phase of the signal  106  is shifted until the signal  106  and the signal  94  are in quadrature at the mixer  110 . Since the signals  92  and  94  preferably differ in phase only based on the phase shift caused by the target sample  32 , each of the signals  92  and  94  are individually placed into quadrature with separate signals in order to determine this difference in phase at the synthesis frequency generator  100 . The phase difference between the signals  104  and  106  thus reflects the phase difference between the input signals  92  and  94 . The fluorescence lifetime of the sample can be calculated using the measured phase shift and frequency with Equation 1. 
   In this embodiment of  FIG. 4 , the frequency generators  82  and  96  are inexpensive, small single integrated circuit, commercially available components. These external generators  82  and  96  do not, however, provide a means for communicating the current phase of the output signals  16  and  26  to the DSP  98 . As a result, the digitized reference signal  92  is at some unknown phase. The additional internal phase lock loop, which is made up of the mixer  108 , the filter  150 , the frequency generator  100  and the frequency calculator  102 , generates a signal  104  that is phase locked to the input signal  92 . The signal  104  then becomes the phase reference for the second mixing process using mixer  110 . 
   The following example illustrates the purpose and function of the additional phase locked loop of FIG.  4 . The signal  92  is digitized at  50  and has some unknown phase which we designate “α”. Simultaneously, the signal  94  is digitized at  48  and differs in phase from the signal  92  by “β”, the result of the phase shift caused by the sensor  32  and any phase shifts due to the analog components  22 ,  36  and  40 . Thus, the signal  94  has a phase of α+β as indicated in FIG.  4 . The phase locked loop which includes the mixer  108  generates the signal  104  with a phase shift of α+90°. The frequency generator  100  then creates a signal  106  that has an added phase shift of “δ” relative to the signal  104 . Therefore, the signal  106  has a phase of α+90 +δ. The mixer  110  and error signal  114  impose on the signal  106  the condition that it must be in quadrature with the signal  94 . This is accomplished by adjusting δ, the amount of additional phase shift relative to signal  104 . At the mixer  110  we find that the signals  94  and  106  differ by 90°, that is the phase of the signal  94  plus 90°, α+β+90, is equal to the phase of signal  106  which we know to be α+90+δ, or
 
α+β+90 =α+90 +δ. 
 
   Simplifying the above, we find that β=δ. The amount of known phase shift added to the signal  106 , δ, is equal to the phase shift caused by the fluorescence experiment  32 , along with the other analog components  22 ,  36 , and  40 . Finally the phase shift indicated by δ may be used with Equation 1 to calculate the fluorescence lifetime. 
   An alternative application of the embodiment of  FIG. 4  includes a frequency feedback signal  154  which passes from the frequency and phase calculator  102  for varying the output of the frequency generators  82  and  96 . In this manner, the frequencies of the signals  16 ,  26  and  84 , and the phase of the signal  106  may also be simultaneously varied as in the embodiment illustrated in FIG.  2 . Another alternative embodiment of  FIG. 4  includes a digital signal processor  98  which contains an internal frequency generator  96  and digital to analog converter  140 . 
   Referring now to  FIG. 5 , this embodiment is similar to that shown in  FIG. 4  except that it lacks a means for downconversion of high frequency signal to lower frequencies for quadrature phase detection. Referring to  FIG. 5 , the down conversion arrangement of  FIG. 4  has been eliminated. This  FIG. 5 , the down particularly useful when only a single phase digital waveform generator is available in place of a dual-phase output digital waveform generator. In one implementation of this embodiment as illustrated in  FIG. 5 , the frequency generator  96  is a specialized DSP component that is external to the main digital signal processor  98  and contains a digital to analog converter  142 . In another implementation, the frequency generator  96  is internal to the main DSP  98 , as shown for example in  FIGS. 1 and 2 . 
   In this embodiment, as in the prior embodiment, the experimental signal  16  and the reference signal  26  are generated by one single-output frequency generator  96  and are generated at substantially the same phase and frequency. As in prior embodiments, the experimental signal  16  passes through the light source  22 , the luminescent sample  32 , and the photodetector  36 . The signal  38  output from the photodetector  36  is converted to a voltage at the preamplifier  40  and filtered at the anti-aliasing filter  42  as in prior embodiments. The experimental output signal  38  is then digitized at the analog-to-digital converter  48 . The reference input signal  26  also passes through a substantially identical anti-aliasing filter  44  and is digitized at the analog-to-digital converter  50 . The digitized representations of the experimental output signal  38  and the reference signal  26  are digital experimental signal  94  and digital reference signal  92 , respectively. 
   As in the prior embodiment of  FIG. 4 , when the signals  92  and  94  pass through the analog-to-digital converters  50 , and  48 , they are not mixed directly together. Instead, an additional dual-output, multiphase digital synthesis frequency generator  100  is provided within the DSP  98 . As in the prior  FIG. 4  embodiment, this generator  100  allows both the digital experimental signal  94  and the digital reference signal  92  to be compared in quadrature at two different digital mixers,  108  and  110 . 
   As in the previously described embodiment of FIG.  4  and in preferred form, the dual-output, multi-phase synthesis frequency generator  100  includes a frequency and phase calculator  102  that generates a first internal signal  104  and a second internal signal  106 , each of which has a frequency which matches substantially exactly the frequency of the signals  92  and  94 , which is the difference frequency between the frequency generated by the generator  96  and the frequency generated by the generator  82 . The signal  104  is generated such that it has a phase of 90° relative to the input signal  92 . This is accomplished by mixing the signals  92  and  104  at a mixer  108 . The output of the mixer  108  is directed to a low pass filter  150  which outputs error signal  152 , the sign and magnitude of which indicates the relative phase difference between signals  104  and  92 . The error signal  152 , preferably zero when the signals  92  and  104  are in quadrature, is directed to the frequency and phase calculator  102 . The calculator  102  adjusts the phase of signal  104  until the error signal  152  indicates that signals  92  and  104  are in quadrature. 
   At the same time, the signal  94  is directed toward a mixer  110 , and generator  100  generates signal  106  of preferably identical frequency with signals  92 ,  94  and  104 , and with phase that is advanced a known amount with respect to the phase of signal  104 . The signal  106  is mixed with the signal  94  at the mixer  110 , the output of which is directed through the low pass filter  56  to create the error signal  114  which is directed to the frequency and phase calculator  102 . Based on the sign and magnitude of the error signal  114 , the phase of the signal  106  is shifted until the signal  106  and the signal  94  are in quadrature at the mixer  110 . Since the signals  92  and  94  preferably differ in phase only based on the phase shift caused by the target sample  32 , each of the signals  92  and  94  are individually placed into quadrature with separate signals in order to determine this difference in phase at the synthesis frequency generator  100 . The phase difference between signals  104  and  106  thus reflects the phase difference between the input signals  92  and  94 . The fluorescence lifetime of the sample can be calculated using the measured phase shift, frequency with Equation 1. 
   As with the prior embodiment, the additional phase locked loop made up of the generator  100 , the feedback signal  152 , the mixer  108  and the integrator  150  allows an additional known amount phase shift to be added to internal reference signal  106  so that the digitized experimental can be compared in quadrature to a signal of known phase. 
   An alternative application of the embodiment of  FIG. 5  includes the frequency feedback signal  154  which passes from the frequency and phase calculator  102  for varying the output frequency of the generator  96 . In this manner, the frequencies of the signals  16  and  26 , and the phase of signal  106  may also be simultaneously varied as in the embodiment illustrated in FIG.  2 . 
   Referring now to  FIG. 6 , this embodiment describes a device  12  that uses a Digital Signal Processor  200  for measuring phase shifts of an analog signal  202  through a phase shifting element  204 , relative to the phase shifts of an analog signal  206  of substantially identical frequency through a reference element  208 . In a preferred embodiment, the analog signal  202  and  206  are preferably generated by a dual-output, variable phase and frequency waveform generator  210 , which is contained in the DSP  200 . The digital outputs,  212  and  214 , of the waveform generator  210  are directed to digital-to-analog converters  216  and  218 . The digital-to-analog converters output analog signals  202  and  206  which are directed to the phase shifting element  204  and the reference element  208 , respectively. 
   In the preferred embodiment of this  FIG. 6 , the phase shifting element  204  contains a fluorescent material that changes lifetime in response to some analyte. Additionally, the phase shifting element  204  may also contain an excitation light source, a photodetector, a pre-amplifier, and anti-aliasing filters as described in the previous embodiments. The reference element  208  preferably contains substantially identical anti-aliasing filters which add substantially the same amount of phase shift to the signal as the anti-aliasing filters of the phase shifting element, as explained in the previous embodiments. It should be noted that the phase shifting element does not necessarily contain a fluorescent sample. It may, in fact, consist of many types of electrical or optical phase shifting components. 
   The output signal  220  of the phase shifting element  204 , and the output signal  222  of the reference element  208  are directed then towards analog-to-digital converters  224  and  226 , respectively. The analog to digital converters  224  and  226  convert the analog signals  220  and  222  into digital representations in the DSP  200 . The digitized signals  220  and  222  are then directed towards a digital phase demodulator  228 . In the preferred embodiment of this  FIG. 6 , the DSP  200  contains a digital phase demodulator  228 , a low pass filter  230 , a dual-output variable-phase waveform generator  210 , and a filtered feedback error signal  232  from the phase demodulator  228 . As described in previous embodiments, these elements act in concert to force the digital signals  220  and  222  into quadrature at the digital phase demodulator  228 . Under conditions of quadrature, the phase shift between the two signals  220  and  222  due to the phase shifting element  204  can be determined using one of the methods of the previous embodiments. 
   This  FIG. 6  embodiment of the invention preferably includes a single analog timing element  234 , which provides a master timing base for all digital signal generation and phase comparison operations in the DSP  200 . In the preferred embodiment, the timing element  234  consists of a quartz crystal oscillator. The timing element  234  generates a high frequency clock signal  236  that id directed to a clock divider  238 . In one preferred embodiment, the high frequency clock signal  236  is approximately 25 MHz. The clock divider  238  then digitally divides the clock signal  236  into a lower frequency clock signal  240 . This lower frequency clock signal  240  becomes the timing signal for all operations relating to the determination of the relative phase between signals  220  and  222 . In the preferred embodiment, the frequency of the clock signal  240  is substantially 48 kHz. Depending on the specific components used, the clock signal  240  may differ significantly from 48 kHz. 
   The clock signal  240  is preferably directed to the digital-to-analog converters  216  and  218  and the analog-to-digital converters  224  and  226 . At the digital-to-analog converters  216  and  218 , the clock signal  240  causes the conversion of a pair of digital points representing waveforms  212  and  214  to analog signals  202  and  206 . Simultaneously, the clock signal  240  causes the analog-to-digital converters  224  and  226  to convert the incoming analog signals  220  and  222  into a pair of digital points representing the analog signals  220  and  222 . Since the analog signal generation and digitization are synchronized to the clock signal  240 , these events occur simultaneously and with essentially no phase jitter. 
   The clock signal  240  additionally causes the phase demodulator  228 , the filter  230  and the waveform generator  210  to perform calculations on the next set of digital numbers  220  and  222 . When the analog signals  220  and  222  are digitized, their digital representations are first directed to a digital phase demodulator  228 . The result of the digital phase demodulator  228  is directed to a digital filter  230 , which outputs a filtered error signal  232  which is then directed to the waveform generator  210 . The waveform generator  210  then generates a new set of digital numbers for digital signals  212  and  214 . The phase and frequency of the digital signals  212  and  214  are determined by the value of the error signal  232 . At every cycle of the clock signal  240 , the above operations are performed once and the operations are completed before the next cycle of clock signal  240 . The waveform generator  210  also provides for a means that the frequency and relative phase of signals  212  and  214  can be output at each cycle of clock signal  240 . Thus, at each cycle of the clock signal  240 , the lifetime of the phase shifting element  204  can be determined essentially continuously using Equation 1. 
   In the preferred embodiment of  FIG. 6 , the clock divider  238 , the digital-to-analog converters  216  and  218  and the analog-to-digital converters  224  and  226  are contained in a integrated circuit separate from the DSP  200 , while the waveform generator  210  is contained within the integrated circuit of the DSP. It should be understood that these components may be contained either within or outside of the DSP  200  without departing from the spirit of the invention. Moreover, the frequencies of the signals  212  and  214  are substantially the same as the frequencies of signals  220  and  222 . It should also be further understood that the phase shifting element  204  and the reference element  208  can include a means for downconverting, as previously described, from the input frequencies  202  and  206  to lower frequencies for signals  220  and  222  without departing from the spirit of the invention. 
   In order to better understand how various of the embodiments of the present invention operate, the following examples are provided. It should be understood, however, that these examples are only for purposes of illustration and are not intended to limit the scope of the invention which is defined by the claims appended hereto. 
   EXAMPLE I 
   The device  10  of  FIG. 1  was implemented using a commercially available ADSP2181 EZLAB prototyping kit from Analog devices and additional analog components as described below. The DSP  12  consisted of an Analog Devices ADSP-2181 KS-133. Dual output waveform generator  14  was implemented in software using Direct Digital Synthesis, a commonly used method for generating digital waveforms (see description in Analog Devices technical specifications for part # AD9830, Rev. A, p. 10). The two digital output signals  16  and  26  were directed to a ΔΣ Stereo (2 channel) CODEC (Analog Devices part # AD1847 JP) which generated two 20 KHz counterpart analog sine-waves with a relative phase difference as specified by the waveform generator  14 . The CODEC output each analog signal with an sampling rate of 48 KHz. 
   One 20 KHz sine wave  16  was directed to an operational amplifier (Analog Devices AD 810 ) that provided sinusoidal current drive to the LED  22 . The light output of the blue LED (Nichia NSPB500S) was immediately filtered using a blue-interference filter that blocked the longer wavelength light (yellow, orange and red) produced by the LED. The resulting blue light,  30 , was directed towards a sample  32 . 
   The sample  32  used in this example and embodiment consisted of platinum-tetrapentafluorophenyl porphyrin (PtTFPP) dispersed in a proprietary oxygen permeable matrix. This sample had a luminescent lifetime of 18.5 microseconds at ambient temperature and pressure. The red luminescence of the sample  34  was directed towards a photodiode  36  having a red interference filter to remove any scattered blue excitation light  30 . The output current  38  of the photodiode  36  (Hamamatsu PIN photodiode S4707-01), was directed towards a transimpedance amplifier  40  (Burr Brown OPA655) with gain which converted the sinusoidally varying photodiode output current into a sinusoidally varying voltage. The voltage signal  38  was directed to an anti-aliasing filter  42 . The anti-aliasing filters  44  and  42  consisted of single-section, low-pass RC filters with time constants of 3.3 μsec. The output of the anti aliasing filter  42  was directed towards the input side  48  of the Stereo CODEC where the signal was sampled and digitized at a rate of 48 KHz. 
   The analog reference signal  26  was directed to an anti-aliasing filter  44 , which consisted of essentially the same components as the anti-aliasing filter  42  in the signal path as described above. The filtered reference signal  46  was directed to the second input  50  of the CODEC and digitized at a rate of 48 KHz, synchronously with the sample analog signal  38 . 
   The digitized representations of the signals  46  and  38  were multiplied point by point at a rate of 48 KHz at phase demodulator  52 . The phase demodulator  52 , implemented in software, multiplied the digitized data pairs of the time series generated by the CODEC  48 ,  50 . The result of the phase demodulator  52  was sent to a low pass filter  56 . The low pass filter  56  consisted of a digital IIR single or double pole low pass filter implemented in the ADSP2181 (see Oppenheim, A. V., and R. W. Schafer.  Discrete - Time Signal Processing.  Englewood Cliffs, N.J.: Prentice-Hall, 1989.) 
   The output  58  of this filter  56 , which represents the sign and magnitude of the phase difference between the signals  46  and  38 , was directed to the dual output waveform generator  14 . The error signal  58  causes the waveform generator  14  to change the phase advance of the reference signal  26  to a value which puts the signals  46  and  38  in quadrature at the phase demodulator  52 . This condition is met when the error signal is zero. 
   Using the sample described above at ambient temperature and pressure, and 20 KHz excitation, the sample produced a phase shift of −21.0°. Several other elements contribute phase shift equally to both the sample channel and the reference channel, (e.g. the CODEC and the anti-aliasing filters) and did not change the relative phase of the two signals. Since there was a −21.9° phase shift though the luminescent sample, the dual output waveform generator  14  adjusted the phase advance of reference signal  26  to 68.1° to achieve quadrature conditions at the phase demodulator  52 . The phase shift of the sample was determined by computing the difference between 90° (quadrature) and the phase advance added to the reference channel, 68.1°, or in other words 90°-68.1°=21.9°. The lifetime of the sample  32  was calculated using Equation 1 above. 
   EXAMPLE II 
   The device  10  of  FIG. 2  was implemented using a commercially available ADSP2181 EZLAB prototyping kit from Analog devices and additional analog components as described below. The DSP  60  consisted of an Analog Devices ADSP-2181 KS-133. Dual output waveform generator  14  was implemented in software using Direct Digital Synthesis as in Example I. The two digital output signals  16  and  26  were directed to a ΔΣ Stereo (2 channel) CODEC (Analog Devices part # AD1847JP) which produced two counterpart analog sine-waves with a relative phase difference and frequency as specified by the waveform generator  14 . For any frequency and phase relationship, the CODEC output the analog signal using a sampling rate of 48 KHz. 
   One sine wave  16  was directed to an operational amplifier (Analog Devices AD810) that provided sinusoidal current drive to the LED  22 . The light output of the blue LED (Nichia NSPB500S) was immediately filtered using a blue-interference filter that blocked the longer wavelength light (yellow, orange and red) produced by the LED. The resulting blue light  30  was directed towards a sample  32 . The sample  32  used in this Example II was similar to that of Example I and consisted of platinum-tetrapentafluorophenyl porphyrin (PtTFPP) dispersed in a proprietary oxygen permeable matrix. However, this sample had a luminescent lifetime of 7.27 microseconds at 100% oxygen at ˜760 Torr and 45° C. 
   The red luminescence  34  of the sample  32  was directed towards a photodiode  36  (Hamamatsu PIN photodiode S4707-01) having a red interference filter to remove any scattered blue excitation light  30 . The output current of the photodiode  36  was directed towards a transimpedence amplifier  40  (Burr Brown OPA655) with gain which converted the sinusoidally varying photodiode output current into a sinusoidally varying voltage. The voltage signal  38  was directed to an anti aliasing filter  42 . The anti-aliasing filters  44  and  42  consisted of single-section, low-pass RC filters with time constants of 3.3 μsec. The output of the anti-aliasing filter  42  was directed towards the input side  48  of the Stereo CODEC where the signal was sampled and digitized at a rate of 48 KHz. 
   The analog reference signal  26  was directed to an anti-aliasing filter  44  which consisted of essentially the same components as the anti-aliasing filter  42  in the signal  38  path. The filtered reference signal  46  was directed to the second input  50  of the CODEC and digitized at a rate of 48 KHz synchronously with the sample analog signal  38 . 
   The digitized representations of the signals  46  and  38  were multiplied point by point at a rate of 48 KHz at the phase demodulator  52 . The phase demodulator  52 , implemented in software, multiplied the digitized data pairs of the time series generated by the CODEC. The resulting signal  54  of the phase demodulator  52  was sent to a low pass filter  56 . The low pass filter consisted of a digital IIR single or double pole low pass filter implemented in the ADSP2181 as previously mentioned for Example I. The output signal  58  of this filter  56 , which represents the sign and magnitude of the phase difference between signals  46  and  38 , was directed to the dual output waveform generator  14 . The error signal  58  caused the waveform generator  14  to simultaneously and continuously change the phase advance of the reference signal  26  and the frequency of signals  16  and  26  to values which puts the signals  46  and  38  in quadrature at the phase demodulator  52 . This condition is met when the error signal is zero. 
   Using the sample  32  described above in 100% oxygen at 760 Torr and 45° C., a 15.940 kHz excitation and phase offset of 53.904° were required to achieve conditions of quadrature at the mixer  52 . Several other elements contributed phase shift equally to both the sample channel and the reference channel, (e.g. the CODEC, and the anti-aliasing filters) and thus did not change the relative phase of the two signals. As previously described, the dual output waveform generator  14  contains a frequency calculator and phase calculator  64  which determines the phase advance of signal  26  and the frequency of signals  16  and  26  based on the feedback error signal  58 . The frequency and phase calculator were set to adjust the phase and frequency according to the following equation:
 
 N°= 0.0038F+6.18   Equation (3) 
 
   The phase advance of the signal  26 , that is N°, and the frequency F of the signals  16  and  26  at any particular phase advance, N°, were determined by Equation 3. The waveform generator  62  increased or decreased N°, and hence simultaneously lowered or raised the frequency F according to Equation 3, until the error signal indicated that the inputs  38  and  46  to the phase demodulator  52  were in quadrature. 
   Since there is a −36.096° phase shift though the luminescent sample, the dual output waveform generator adjusted the phase advance of reference signal  26  to 53.904° to achieve quadrature conditions at the phase demodulator  52 . Following Equation 3, the frequency was set to 15.94 kHz. The phase shift of the sample was determined by computing the difference between the phase advance added to the reference channel, 53.904°, and 90° (quadrature), or 53.904°−90°=−36.096°. The lifetime of the sample then was calculated using Equation 1 above. 
   EXAMPLE III 
   The device  10  of  FIG. 3  is implemented using a commercially available ADSP2181 EZLAB prototyping kit from Analog devices and additional analog components as described below. The DSP  60  consists of an Analog Devices ADSP-2181 KS-33. Dual output waveform generator  62  is implemented in software using Direct Digital Synthesis as previously mentioned in the Examples I and II, or if the output frequencies required are too high for generation in software, the waveform generator is implemented externally in a custom DSP chip and clocked by the same master analog clock as the other digital components. The waveform generator  62  generates two digital output signals  16  and  26  that are then directed to an appropriate digital-to-analog converter  20  which produces two counterpart analog sine-waves with a relative phase difference and frequency as specified by the waveform generator  62 . 
   As in the previous examples, one sine wave  16  is directed to an operational amplifier (analog Devices AD810) to provide sinusoidal current drive to the LED  22 . The light output of a blue LED (Nichia NSPB500S) is immediately filtered using a blue-interference filter that blocks the longer wavelength light (Yellow, orange and red) produced by the LED. The resulting blue light  30  is then directed towards a sample  32 . The sample  32  consists of a fluorescent sample the lifetime of which is quenched in the presence of the analyte of interest. The lifetime of the sample can be quite short, for example 0.5 nsec to 5 nsec. At 10 MHz excitation, a fluorescent sample with a 5 nsec lifetime exhibits a phase shift of 17°. 
   The longer wavelength fluorescence of the sample  34  is directed towards a photodetector  36 , such as a photomultiplier tube, avalanche photodiode or other appropriate detector, having an appropriate interference filter to remove any scattered blue excitation light  30 . The output current  38  of the photodetector  36  is directed towards a transimpedance amplifier  40  (e.g. Burr Brown OPA655) to convert the sinusoidally varying photodetector output current into a sinusoidally varying voltage  38 . The voltage signal  38  is then directed to an analog mixer  90 . The analog reference signal  26  is also directed to similar analog mixer  88 . 
   This embodiment provides for a second waveform generator  82 , which in this example is a commercially available AD9830. This waveform generator  82  outputs a high frequency digital signal  84  which is then converted into an analog signal at an appropriate digital-to-analog converter  86  as previously described. The digital-to-analog converter  86  may in fact be integral to the second waveform generator  82 . The second waveform generator  82  outputs a signal  84  with a frequency that is a constant difference from the frequencies of signals  16  and  26 . For example, if the signals  16  and  26  are 10 MHz signals, the second waveform generator  82  outputs a signal of 10.02 MHz. In this example, the constant difference between the first generator  62  and second generator  82  is 20 KHz. If the output frequency of the waveform generator  62  changes, the output of the second generator  82  tracks the output frequency of the first generator  62  such that the difference in frequencies remains 20 kHz. 
   The analog output  84  of the second generator is split into two identical signals directed towards the analog mixers  88  and  90 . Both analog mixes  88  and  90  perform substantially identically, their outputs being a linear multiplication of the two input signals. These outputs  92  and  94  contain the sum and difference frequencies of the input signals to the respective mixers  88  and  90 . In this example, the mixer output thus consists of a 20.02 MHz signal and a 20 kHz signal. 
   The sum and difference frequency outputs of the analog mixers  88  and  90  are filtered at similar anti-aliasing filters  44  and  42 . These anti-aliasing filters  44 ,  42  are configured so that the sum frequencies, 20.02 MHz, are removed, leaving only the difference frequencies, i.e. 20 kHz. As described above, the downconverted 20 kHz difference frequencies carry the same relative phase information as did the 10 MHz signals that passed through the fluorescent sample and the reference path. Appropriate anti-aliasing filters are, for example, single-section, low-pass RC filters with time constants of 3.3 μsec. 
   The output signal of each of the anti-aliasing filters  42  and  44  is directed towards the input of an analog-to-digital converter  48 ,  50 , respectively, where the signal is sampled and digitized at an appropriate rate, e.g. 48 KHz. The digitized representations of filtered signals  92  and  94  are then multiplied point by point at a rate of 48 KHz at the phase demodulator  52 . The phase demodulator  52 , implemented in software, multiplies the digitized data pairs of the time series generated by the analog-to-digital converter. The resulting signal  54  of the phase demodulator  52  is sent to a low pass filter  56 . The low pass filter consists of a digital IIR single or double pole low pass filter implemented in the ADSP2181 as previously mentioned. 
   The output signal  58  of this filter  56 , which represents the sign and magnitude of the phase difference between filtered signals  92  and  94 , is directed to the high-frequency dual-output waveform generator  62  as well as the second high frequency generator  82 . The error signal  58  causes the waveform generator  62  to simultaneously and continuously change the phase advance of the reference signal  26  and the frequency of signals  16  and  26  to values which puts the filtered signals  92  and  94  in quadrature at the phase demodulator  52 . This condition is met when the error signal is zero. 
   The dual output waveform generator  62  can be operated in one of two modes. It can be operated in a first mode at a constant frequency and variable phase shift, as in Example I, or it can be operated in a second mode simultaneously at a continuously variable frequency and phase with compression, as in Example II. If the waveform generator  62  is operated in the second mode, the error signal  58  also acts on the second high frequency generator  82  to adjust its frequency so that it is at a constant 20 kHz difference from the frequency output of the first generator  62 . The sample lifetime is calculated as in Examples I and II using the phase shift and frequency with Equation 1. 
   EXAMPLE IV 
   The device  10  of  FIG. 4  was implemented using a commercially available ADSP2181 EZLAB prototyping kit from Analog devices and additional analog components as described below. The DSP  98  consisted of an Analog Devices ADSP-2181 KS-133. Unless stated otherwise, the components of this Example IV were the same specific components utilized in the previous examples. Instead of the dual-output waveform generator described in the Example III above, a single-output Direct Digital Synthesis waveform generator  96  was used to generate a high frequency sinusoid that drove an LED  22  to generate blue light for excitation of the sample. The waveform generator  98  consisted of an Analog Devices EVAL-AD9830EB which contained an analog Devices AS9830 Direct Digital Synthesis IC. 
   The waveform generator  96  contained an internal analog-to-digital converter  142  and output a sine wave of a known frequency, i.e. 1.013 MHz. This sine wave output was split into two identical signals  16  and  26 . One signal  16  was directed towards an operational amplifier, Analog Devices AD811, which provided sinusoidal current drive to a blue LED  22  (Nichia NSPB500S), and the other signal, the analog reference signal  26 , was sent to an analog mixer  88 . 
   The light output of a blue LED  22  (Nichia NSPB500S) was immediately filtered using a blue-interference filter that blocked the longer wavelength light (yellow, orange and red) produced by the LED. The resulting blue light  30  was directed towards a sample  32 . The sample  32  consisted of a dilute fluorescent sample in a buffer of pH 7.6. The lifetime of the sample was 4 nsec under ambient conditions. 
   The longer wavelength luminescence of the sample  34  was directed towards a photomultiplier tube  36  (Hamamatsu R5600U-01) which has a 600 nm longpass filter to remove any scattered blue excitation light  30 . The output current  38  of the photomultiplier tube  36  was directed towards a transimpedance amplifier  40  (e.g. Burr Brown OPA655) to convert the sinusoidally varying photodetector output current into a sinusoidally varying voltage. The voltage signal  38  was directed to an analog mixer  90 . The analog reference signal  26  was directed to a similar analog mixer  88 . 
   This device  10  had a second waveform generator  82 , which consisted of a Analog Devices EVAL-AD9830EB containing an analog Devices AS9830 Direct Digital Synthesis IC. This second waveform generator  82  output a high frequency analog signal  84  that was a constant difference from the frequencies of signals  16  and  26 . In this example the second oscillator frequency was 1.0078 MHz, a constant difference from the first oscillator frequency of 5.2 kHz. The analog output  84  of the second generator  82  was split into two identical signals directed towards the analog mixers  88  and  90 . Both analog mixers  88 ,  90  acted substantially identically in that the output of each was a linear multiplication of their two respective input signals, i.e. signals  26  and  84  for the mixer  88  and signals  38  and  84  for the mixer  90 . These outputs  92  and  94  contained the sum and difference frequencies of the input signals. In this example, the mixer output consisted of a 2.0208 MHz signal and a 5.2 kHz signal. 
   The sum and difference frequency outputs of the analog mixers  88 ,  90  were filtered at similar anti-aliasing filters  44  and  42 . These anti-aliasing filters were configured so that the sum frequencies, 2.0208 MHz, were removed, and the difference frequencies, i.e. 5.2 kHz, were passed on. As described above, the downconverted 5.2 kHz difference frequencies carried the same relative phase information as did the 1.013 MHz signals  16 ,  26  that passed through the luminescent sample  32  and that acted as the analog reference signal, respectively. Single-section, low-pass RC filters with time constants of 3.3 μsec were used to filter the downconverted signals. 
   The output of the anti-aliasing filters  44  and  42  were directed to the inputs  50 ,  48  of a ΔΣ Stereo (2 channel) CODEC (Analog Devices part # AD1847 JP) where the signals were digitized at a rate of 48 KHz. The digitized filtered reference signal  92  was directed to a digital phase demodulator  108 . A second input to the digital phase demodulator was a 5.2 kHz digital sine wave  104 , generated by an internal dual-output waveform generator  100 . The digital phase demodulator  108  consisted of a point by point multiplication which operated at 48 kHz, which is the rate of digitization of the CODEC. 
   The error signal generated by this phase demodulator  108  was then filtered with a digital IIR single or double pole low pass filter  150  implemented in the ADSP2181 as previously mentioned. The filtered error signal  152  was input to the dual-output waveform generator  100 , which adjusted the phase of the digital sine wave  104  such that it was in quadrature with the digitized reference signal  92  at the demodulator  108 . Because the reference signal  92  has an unknown phase of α, as noted in  FIG. 4 , the digital phase lock loop described above locks one output,  104 , of the internal dual-output waveform generator to the digitized reference input, signal  92 . Specifically, the action of the mixer  108  and the feedback error signal  152  causes signal  104  to have a phase shift 90° advanced with respect to the signal  92 , or α+90°. 
   The 5.2 kHz downconverted digitized signal from the fluorescence experiment, signal  94 , was directed towards a second digital phase demodulator  110  in the DSP. The second input to this phase demodulator was the second output  106  of the internal dual-output waveform generator  100 . As with the other digital phase demodulator  108 , the result of the point-by-point multiplication passes through a IIR single or double pole low pass digital filter  56 . This filtered signal  114 , the error signal of the phase demodulator  110 , was directed towards the phase and frequency calculator  102  of the internal dual-output waveform generator  100 . The phase and frequency calculator  102  caused the waveform generator  100  to output a signal  106  at 5.2 kHz with a phase 90° advanced with respect to the digitized signal from the fluorescence experiment, signal  94 . 
   Because both signals  104  and  106  generated by the internal dual-output waveform generator  100  are in quadrature with the downconverted digitized reference and fluorescence experiment signals  92  and  94 , respectively, the phase difference between the two signals generated by the internal dual-output waveform generator  100  reflect the phase difference between the digitized reference and fluorescence experiment signals  92 ,  94 . Thus, the total phase delay resulting from the contribution of the excitation light source, the fluorescence experiment, the photodetector and the transimpedance amplifier are known. Since the phase delay added by the fluorescence experiment  32  is the parameter of interest, the phase delays due to other components of the system were factored out by measuring a fluorescence sample of known lifetime. 
   In this particular example, the phase delays of the excitation light source, fluorescent sample (fluorescein), the photodetector and the pre-amplifier amounted to −13.42°. This was measured by determining the difference between the two digital output signals  104 ,  106  of the internal dual-output waveform generator  100 , while both of the phase demodulators  108 ,  110  in the DSP were operating in quadrature. Of the −13.42° phase shift, −1.46° were due to the 4 ns lifetime of the fluorescent sample, and −11.96° resulted from phase shift in the excitation, photodetector and trans-impedance amplifier. The non-fluorescence sample phase shifts of −11.96° were assumed to be constant and thus subtracted from the measured phase difference between the downconverted signals  92  and  94 . Equation 1 was used with θ=−1.46°, and ƒ=1.013 MHz to calculate the fluorescence lifetime of the sample. 
   As can be seen from the above, it is clear that the present invention provides a simple and effective apparatus and method for measuring environmentally and medically important analytes. The present invention accomplishes this by providing a unique apparatus and method for measuring time delay in samples targeted by electromagnetic radiation and in particular fluorescence emissions. The present invention provides a fluorescence-based sensing instrument and method applicable to a broad range of materials involving exponential decay and time delay measurements, and which is made from inexpensive components. While analog systems of the present art are subject to drift and therefore unnecessary errors, and the digital systems of the present art contain complex, expensive hardware, the present invention has been designed to avoid these problems. Consequently, the system of the invention is inexpensive, convenient to use and operates over an extended and continuous measurement range. In addition, the measurement approach of the device and method of the invention is susceptible to convenient and precise readout. 
   The foregoing description and the illustrative embodiments of the present invention have been described in detail in varying modifications and alternate embodiments. It should be understood, however, that the foregoing description of the present invention is exemplary only, and that the scope of the present invention is to be limited to the claims as interpreted in view of the prior art. Moreover, the invention illustratively disclosed herein suitably may be practiced in the absence of any element which is not specifically disclosed herein.