Abstract:
A circuit for providing a secondary output voltage from an input voltage. The circuit comprises power supply circuitry for creating an unregulated DC bus voltage line, a regulator circuit connected to the DC bus voltage line for controlling a first switch in series with a transformer winding, the control circuit sampling an output voltage to control the output voltage by cycling the switch, a pulse generator circuit connected to the regulator circuit for controlling start and stop cycles of the regulator circuit, and a comparator circuit connected to the pulse generator circuit, for monitoring the secondary output voltage and disabling the pulse generator circuit during normal operation of the power supply circuit.

Description:
FIELD OF THE INVENTION  
         [0001]    The present invention relates to a power supply start up circuit and more particularly to a power supply start up circuit designed to inhibit power losses by dissipation therefrom.  
         BACKGROUND OF THE INVENTION  
         [0002]    Switching power supplies of various topologies are used to provide a regulated output voltage (Vreg) from an unregulated or regulated input voltage (Vin).  
           [0003]    As discussed in greater detail below, prior art switching power supplies suffer from prolonged or erratic start-up times as a result of the time constraints in standard prior art resistor capacitor startup circuits.  
           [0004]    Also, prior art switching power supplies are prone to continuous power dissipation in the bleeder resistor of the standard resistor capacitor startup circuit during normal operation.  
           [0005]    Furthermore, prior art switching power supplies suffer from excessive heating of circuit components during overload or short circuit conditions where such heating is as a result of the output being cycled on and off at a rate determined principally by the time constraints in the standard resistor capacitor startup circuit. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0006]    The invention and the prior art will be better understood with reference to the drawings and the following description in which:  
         [0007]    [0007]FIG. 1 is a schematic diagram of a typical industry standard switching power supply circuit of the prior art;  
         [0008]    [0008]FIG. 2 is a schematic diagram of a power supply start up circuit according to an embodiment of the present invention;  
         [0009]    [0009]FIG. 3 is a schematic diagram of a power supply start up circuit according to a second embodiment of the present invention; and  
         [0010]    [0010]FIG. 4 is a schematic diagram of a power supply start up circuit according to a third embodiment of the present invention. 
     
    
     DETAILED DESCRIPTION OF THE PRIOR ART  
       [0011]    [0011]FIG. 1 is a schematic diagram of a typical industry standard switching power supply circuit of the prior art, commonly referred to as a “flyback” topology. Referring to FIG. 1, a commercial AC input voltage (Vin) is stepped down by a transformer (wall adapter power supply) Twa. The wall adapter power supply is connected to an AC/DC rectifier diode D 1  that is in turn connected to a capacitor C 1  to create an unregulated DC bus voltage (Vbus).  
         [0012]    A control circuit U 1  controls the operation of a transistor switch Q 1  that is connected in series to a primary transformer winding T 1 A. The control circuit U 1  controls secondary output voltages Vaux and Vreg by varying on and off times of the transistor switch Q 1 . In the present example of the prior art, the output voltage Vreg is isolated by a standard optocoupler Ufbk such that a regulated voltage is maintained at various values of external load resistance Rload and input power Vin.  
         [0013]    During the initial application of input power Vin, the capacitor C 1  charges up to the peak value of the rectified AC input voltage, (or DC input voltage value if the power supply is powered by a DC input). A small charging current flows from the bus voltage Vbus through the resistor R 1  and into the capacitor C 2  causing its voltage to rise. When the voltage across the capacitor C 2  exceeds the start threshold value for the control circuit U 1  (typically 16 VDC), transistor switch Q 1  is switched on causing AC current to flow in the primary transformer winding T 1 A. The secondary output voltages Vaux, Vreg are then induced by transformer action in output windings T 1 B and TIC causing the secondary output voltages Vreg, Vaux to rise to steady state values. While the secondary output voltages Vreg,Vaux are rising, the voltage across the capacitor C 2  is steadily decreasing due to the fact that the operating current of the circuit U 1  is typically more than can be supplied through the resistor R 1  alone. For sustained operation, the output voltage Vaux must rise quickly enough to prevent the capacitor C 2  from dropping below the control circuit U 1  stop threshold voltage. If this does not occur, the output voltage Vaux will drop below the stop threshold of the control circuit U 1  (typically 10 VDC) and the power supply secondary output voltages will drop to zero. A new start cycle must then be initiated and the process is repeated until the power supply starts.  
         [0014]    The prior art switching power supply of FIG. 1 suffers from a number of disadvantages. Firstly, the value of the resistor R 1  is typically chosen as a compromise between low power dissipation and reliable startup characteristics under low input power conditions (Vin) and high current external load (Rload) conditions. Since the input voltage Vin can typically vary between 20 VDC and 33 VDC while the output voltage Vaux is typically approximately 12 VDC, the resistor R 1  is usually chosen to have a high resistance value in order to minimise the current flowing through it and subsequent power dissipation within it. This is desirable, as power loss in the resistor R 1  reduces the efficiency of the power supply and increases internal heating, thereby reducing the reliability of the power supply. At high DC bus voltages (Vbus) this problem becomes more severe as the power dissipated in the resistor R 1  increases as the square of voltage (Power=Vbus×Vbus/R 1 ). If, however, the resistance of the resistor R 1  is too high, the charging current available through it may be insufficient to overcome the combined leakage current of the capacitor C 2  and the control circuit U 1 . Especially at low values of DC bus voltage (Vbus)this can result in the voltage across the capacitor C 2  failing to reach a value equivalent to the start threshold voltage of the control circuit U 1  and consequent failure of the power supply to start.  
         [0015]    An additional problem with this prior art circuit is that, after the supply is operating normally and the control circuit U 1  is being powered principally from the secondary output voltage Vaux, charging current still flows through the resistor R 1  from the bus voltage (Vbus) to the output voltage (Vaux). Power is therefore dissipated within the resistor R 1  continuously. At high bus voltage (Vbus) values the dissipation increases as the square of the bus voltage, (Vbus) as described above. This means that the resistor R 1  must be sized for continuous dissipation at these worst case conditions, thereby increasing the size and cost of the resistor R 1 . This results in internal heating that reduces the lifetime of other components in the vicinity of RI as well as lowering the efficiency and increasing the operating cost of the power supply.  
         [0016]    Another problem with the prior art circuit of FIG. 1 is that the value of the capacitor C 2  is typically chosen to provide sufficient energy storage to allow the control circuit U 1  to continue operating long enough for the output voltages Vreg, Vaux to reach their steady state values as explained above. If the value of the capacitor C 2  is too small it may not retain sufficient energy to enable the power supply to start before the voltage across it drops below the control circuit U 1  shutdown threshold voltage. This is a problem with high current external load (Rload) values which may require several switching cycles to build up the output voltage Vreg and subsequently the output voltage Vaux to their steady state values. One solution is to make the value of the capacitor C 2  larger in order to have more energy storage available for a longer control circuit U 1  operating time. However, if the capacitor C 2  is made too large without changing the resistor R 1 , more time is required to charge C 2  up to the control circuit U 1  start threshold voltage value. This is especially true under conditions of low bus voltage (Vbus) when the resistor R 1  charging current is lowest. This can result in excessively long power supply start up times which are inconvenient and generally cause confusion in the mind of the user as to whether the unit is defective or not. Attempts to remedy this situation by changing the value of the resistor R 1  result in all of the problems related to the selection of this component described above.  
         [0017]    Yet another disadvantage of this prior art circuit is that the resistor R 1  and the capacitor C 2  in combination determine the time interval between successive start and restart cycles such as occur when the power supply is in what is called, “current limit” mode. Current limit occurs when the resistance of the external load Rload drops to a very low value such that the current flowing through it increases beyond the design limit of the power supply. In typical switching power supply circuits the control circuit U 1  senses this condition by monitoring the voltage drop across resistor R 2 , although other types of current sensing may also be used. When the voltage across the resistor R 2  exceeds a minimum threshold, the control circuit U 1  acts upon the ON time of the transistor switch Q 1  to reduce the output voltage Vreg. As the load current increases, the output voltage Vreg is further reduced. As the output voltage Vreg drops, the output voltage Vaux is also reduced by the transformer action of the windings T 1 B, T 1 C. When the output voltage Vaux drops to less than the control circuit U 1  stop threshold, the transistor switch Q 1  stops switching, causing the output voltages Vreg, Vaux to drop to zero. The capacitor C 2  then begins to recharge through the resistor R 1  until its voltage reaches the control circuit U 1  start threshold, whereupon the power supply attempts to restart. These current limit shutdown and restart cycles repeat until the overload is removed. The problem here is that the interval between restarts is determined by how quickly the capacitor C 2  can charge from the stop to start thresholds of the control circuit U 1 . This “cycle” time is therefore much less than the start time since the capacitor C 2 , in the example of FIG. 1, only has to charge up from 10 VDC to 16 VDC compared to charging from 0 to 16 VDC as under normal start conditions. This “cycle” time is further reduced as the input voltage Vin increases due to greater charging current through the resistor R 1 . The net effect of decreasing the time interval between the stop and start cycles during conditions of current limit overload is to increase the heating and electrical stress of the transistor switch Q 1 , the transformer coils T 1 A, T 1 B, T 1 C, the diodes D 1  and D 3  as well as other current carrying printed wiring board or wiring paths and power connectors within the power supply. These stresses reduce the reliability and operating life of the power supply.  
         [0018]    To size these components to operate at reduced temperatures under overload conditions would increase the cost and physical space required by them. Depending on the requirements of products in which the power supply is intended to be incorporated this may not be an option. Another solution is to extend this time interval between the stop and start cycles during current limit overload conditions. One method of doing this is to increase the voltage range between the capacitor U 1  stop and start thresholds. Unfortunately most commercially available controller IC&#39;s have fixed stop and start threshold voltages that cannot be adjusted. Both these and discrete circuits are also typically limited by the minimum and maximum drive voltages required for the transistor switch Q 1  especially if this device is a MOSFET power transistor (i.e. industry standard in this application). Yet another solution is to increase the values of the resistor R 1  and the capacitor C 2  to create a longer delay between stop and start cycles. Unfortunately, this has the unwanted effect of delaying the power supply startup time as described above. Also, increasing the value of the capacitor C 2  will increase the time during which the power supply operates in the overload condition thereby increasing electrical stress and thermal dissipation in the unit.  
       SUMMARY OF THE INVENTION  
       [0019]    As stated above, the typical startup circuit shown in FIG. 1 suffers from many disadvantages.  
         [0020]    The value of the resistor R 1  must be a compromise value. This value must be low enough to provide sufficient charging current to the capacitor C 2  to ensure reliable startup in a reasonable period of time at low Vin conditions but high enough to minimise power dissipation of the resistor R 1  at high Vin conditions.  
         [0021]    The value of the capacitor C 2  must also be a compromise. It must be high enough to provide sufficient energy storage for startup under full load conditions at low Vin but low enough to avoid excessively long startup times under the same conditions.  
         [0022]    Furthermore, the resistor R 1  and capacitor C 2  in combination must satisfy the above two conditions as well as provide a stop/start cycling time interval during current limit overload conditions which limits the heating and electrical stress in the current carrying components within the power supply.  
         [0023]    Since there are conflicting requirements for the optimisation of the resistor R 1  and capacitor C 2 , practical circuits using the startup configuration of FIG. 1 typically compromise between power supply performance and reliability. For example, the power supply may start up under maximum load in a reasonable time period but may require several tries to start under these conditions at low Vin. Also, the power supply may start in a reasonable period of time but may suffer from excessive dissipation in RI during operation at high Vin, thereby reducing the power supply service life. Further, the power supply may start quickly under low Vin conditions but may fail during sustained operation during overload conditions due to rapid start/stop cycling and resultant overheating.  
         [0024]    It is an object of the present invention to obviate or mitigate at least some of the disadvantages of the prior art.  
         [0025]    In one aspect of the present invention, there is provided a circuit for providing a secondary output voltage from an input voltage. The circuit comprises power supply circuitry for creating an unregulated DC bus voltage line, a regulator circuit connected to the DC bus voltage line for controlling a first switch in series with a transformer winding, the control circuit sampling an output voltage to control the output voltage by cycling the switch, a pulse generator circuit connected to the regulator circuit for controlling start and stop cycles of the regulator circuit, and a comparator circuit connected to the pulse generator circuit, for monitoring the secondary output voltage and disabling the pulse generator circuit during normal operation of the power supply circuit.  
       DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT  
       [0026]    Reference is made to FIG. 2 to describe a first embodiment of a power supply start up circuit according to the present invention. AC input power (Vin) is connected to a transformer (wall adapter power supply) Twa. The transformer is connected to an AC/DC rectifier diode D 1  which, in turn, is connected to a capacitor C 1  to create an unregulated DC bus voltage Vbus.  
         [0027]    A pulse generator circuit is used to control the operation of a transistor switch Q 2 . The pulse generator circuit includes an open collector comparator U 2 , input resistors R 3 , R 4  and a feedback resistor R 12 . A second comparator U 3  is used to control the first comparator U 2  to prevent repeated cycling, as will be explained further below.  
         [0028]    When Vin is first applied, capacitor C 1  is charged through AC/DC rectifier diode D 1 . The DC bus voltage Vbus is divided across resistors R 3 , R 4  and connected to the comparator U 2  (+).  
         [0029]    The voltage at this point is altered by the state of the output of the comparator U 2  through the feedback resistor R 12 . When the output of the comparator U 2  is high, the transistor switch Q 2  is off and the U 2 (+) is shifted high as resistor R 12  is effectively in parallel with resistor R 3 . When the output of the comparator U 2  is low, Q 2  is on and U 2 (+) is, in turn, shifted low as now R 12  is effectively in parallel with R 4 . This provides alternating high and low offset threshold voltages at U 2 (+). From the high state of comparator U 2  with switch Q 2  off, capacitor C 5  charges through resistor R 5 , diode D 7 , resistor R 7 , and resistor R 8  causing the U 2 (−) voltage to rise. When the U 2 (−) voltage exceeds the U 2 (+) voltage, the output of the comparator U 2  switches to the low state, turning on the switch Q 2  and offsetting the U 2 (+) threshold to a low value. The capacitor C 5  then discharges through resistor R 6  and diode D 6  until the voltage at U 2 (−)drops below the U 2 (+) low threshold at which point the output of comparator U 2  switches to its former high state, thereby turning off the switch Q 2 . The circuit continues to cycle the switch Q 2  on and off as the capacitor C 5  is charged and discharged. The value of C 5  and the resistors R 5 , R 7  and R 8  determine the time constants for on and off time intervals of switch Q 2 , (also referred to as start and stop cycle time). In choosing the value of the capacitor C 5  charge time constant (Q 2  off time), the total resistance value is made high to reduce the current drawn through resistor R 7  from Vbus and thereby prevent inadvertently turning on switch Q 2 .  
         [0030]    When switch Q 2  turns on, resistor R 1  is effectively connected to the DC bus voltage Vbus and starts charging capacitor C 2 . When the voltage across the capacitor C 2  exceeds the start threshold voltage of control circuit U 1 , switch Q 1  is enabled and the output voltages Vreg, Vaux increase to their design values. When this occurs, sufficient voltage is available across transformer coil TIC to keep capacitor C 2  charged through diodes D 2 , D 5  for continuous operation.  
         [0031]    The start cycle time for the pulse generator circuit is set by the discharge time constant of capacitor C 5  and resistor R 6 . This is made longer than the power supply start time set by the R 1  C 2  time constant at low input voltage Vin, when the available capacitor C 2  charging current is low. The start cycle time can also be further increased should additional start time be required due to high starting loads such as occur with capacitive or reactive output loads.  
         [0032]    When the voltage across capacitor C 5  or U 2 (−) drops below that of U 2 (+), the output of the comparator U 2  switches to a high output level. Thus, switch Q 2  turns off and resistor R 1  is disconnected from Vbus. In this manner the dissipation of resistor R 1  is reduced to zero. Because resistor R 1  only dissipates power for a short time period, (i.e. the start cycle time), the resistor R 1  can be optimised to quickly charge capacitor C 2  with sufficient energy to provide reliable startup under heavy Rload conditions at low input voltage Vin. Also, the continuous power rating and hence physical size and cost of resistor R 1  is minimised.  
         [0033]    In order to prevent comparator U 2  from cycling switch Q 2  on and off repeatedly the comparator U 3  is used to disable comparator U 2  and keep switch Q 2  off during normal operation. Comparator U 3  does this by monitoring the voltage across capacitor C 4  and comparing it to a reference voltage derived from the forward voltage of diode D 4 . Diode D 5  isolates the monitored voltage from the voltage across capacitor C 2  to inhibit erratic operation during startup. Capacitor C 4  provides local filtering of the rectified voltage from transformer coil TIC. When this voltage reaches its normal operating level the voltage across the R 9 /R 10  divider seen at U 3 (−) exceeds that of U 3 (+) causing the output of comparator U 3  to switch to a low voltage level, thereby discharging capacitor C 5 . Comparator U 2  is thus disabled with its output in the high or non-conducting state. Switch Q 2  is, by extension, turned off and no current flows through resistor R 1 .  
         [0034]    While the transformer T 1 C (Vaux) voltage is monitored in FIG. 2, Vreg or any other secondary output could be monitored to disable the pulse generator circuit, as would occur to those of skill in the art. If Vreg is an isolated output, as shown in FIG. 1, an isolating device similar to the optocoupler shown in the Ufbk block could be used to provide the required isolation. If a current limit overload should occur on output voltage Vreg, the resulting switch Q 1  current is sensed across resistor R 2  by control circuit U 1  whereupon the switching action of switch Q 1  is reduced such that output voltage Vreg is decreased with increasing load. As this load increases, the voltage across transformer coil T 1 C drops and at some point will fall below the stop threshold of control circuit U 1  causing switch Q 1  to cease switching and output voltage Vreg to drop to zero. When this occurs, the output of comparator U 3  goes high allowing capacitor C 5  to resume its charge/discharge cycle turning switch Q 2  off and on as described above. If the output voltages Vreg, Vaux do not reach their nominal values as a result of the overload by the time the start cycle terminates and the stop cycle begins (i.e. a time interval determined by the R 6 /C 5  discharge time constant), the start and stop cycles are repeated until the overload is removed.  
         [0035]    Diodes D 6  and D 7  provide separate discharge and charge paths for C 5  thereby allowing the start and stop cycles to have different time intervals. It is desirable to make the stop time longer than the start time to reduce the dissipation during sustained overload operation.  
         [0036]    The current required by this circuit and subsequent power loss within it should be considered in component selection, as would occur to those of skill in the art. Because the circuit requires very low current to operate, and the required current is further reduced when the circuit is disabled during normal operation of the power supply, the net power loss is reduced compared to that required by the circuit shown in FIG. 1.  
         [0037]    Reference is now made to FIG. 3 to describe a second embodiment of a power supply start up circuit according to the present invention. The present embodiment is similar to the first embodiment of FIG. 2, however, an additional transistor switch (Q 3 ), diode (D 8 ) and resistor (R 13 ) combination is added to the Q 2  pulse generator circuit to provide a substantially constant capacitor C 2  charging current. This results in a substantially constant startup time for the power supply over a range of DC bus voltages Vbus.  
         [0038]    In operation, when the Q 2  collector current reaches a threshold, set by the point at which the voltage drop across resistor R 13  exceeds the base-emitter voltage of Q 3  plus the forward drop across diode D 8 , switch Q 3  turns on. The effect of this is to divert base drive current away from switch Q 2  causing it to turn off. This results in an increased voltage drop across the Q 2  collector-emitter junction with less current flowing through R 1 . As this current falls, switch Q 3  turns off as the base-emitter voltage of Q 3  plus forward drop across diode D 8  exceeds the voltage drop across R 13 . Thus, the current through R 1  remains substantially constant. Diode D 8  is required to cancel out the voltage drop across the Q 2  base-emitterjunction.  
         [0039]    Referring now to FIG. 4, a third embodiment of the present invention is shown in which the circuit can be used. The circuits of FIGS. 2 and 3 are powered from a low voltage wall adapter power supply Twa. In the third embodiment, the wall adapter power supply Twa is replaced by a direct connection to the input voltage Vin. As in the previous embodiments, a diode D 1  is used to provide an unregulated DC bus voltage Vbus.  
         [0040]    The typical Vbus voltage range corresponds roughly to the peak AC value of Vin. For a universal input switching power supply, known to those of skill in the art, Vbus can typically vary from about 125 VDC to 370 VDC over the 90 VAC to 264 VAC range. Since the industry standard comparators U 2  and U 3  typically have a voltage rating of less than 40 VDC, the present embodiment includes voltage level shifting transistors and other modifications to the first embodiment, to enable operation with a high DC bus voltage Vbus.  
         [0041]    In operation, when the input voltage Vin is first applied the DC bus voltage Vbus rises to its steady state value. Current flows through resistor R 14  and turns on switch Q 4  which, in turn, turns on switch Q 3 , thereby connecting the DC bus voltage Vbus to resistor R 11 . The resistor R 11  and regulating diode D 8  maintain a substantially uniform voltage across comparator U 2  (typically 12 VDC) over the DC bus voltage range described above. Comparator U 2  operates in the same manner as in the above-described embodiments with the exception that level shifting transistors Q 5 , Q 6  and associated resistors R 15 , R 16  are added to keep the U 2  output transistor isolated from DC bus (Vbus) voltage levels. When the power supply begins its start cycle the voltages across the secondary outputs rise. When the voltage across capacitor C 4  reaches its nominal voltage, the level at U 3 (−) set by R 9 /R 10 , exceeds the reference voltage set by the regulating diode D 6  at U 3 (+). The comparator U 3  output then switches to a lower level, causing switches Q 4  and Q 3  to turn off. In the present embodiment, capacitor C 6  is added to increase the time delay of the switching action of comparator U 3  to make the circuit insensitive to transient voltage conditions such as may arise during the start cycle or even during normal operation.  
         [0042]    Diode D 9  ensures that capacitor C 6  is quickly discharged during a stop cycle to inhibit erratic operation during subsequent start cycles caused by residual voltage across capacitor C 6 . When switch Q 3  turns off, the comparator U 2  start circuit is disabled and power dissipation in resistors R 11 , R 1  is reduced. The remaining power dissipated by the startup circuit from the DC bus voltage Vbus is confined to resistor R 14  through comparator U 3 . Since resistor R 14  provides only a fraction of a milliamp to operate switch Q 4 , the total power dissipation is very low.  
         [0043]    Alternative embodiments and variations of the invention are possible, for example, a single diode D 1  is shown but any standard diode and capacitor configuration can be used to provide an unregulated DC output voltage (Vbus) from the AC input. Also, although the output voltage Vreg is isolated by the optocoupler, Vreg need not necessarily be isolated. While a comparator circuit is shown, any industry standard equivalent pulse generator circuit can be substituted as would occur to those of skill in the art. Other variations and modifications would occur to those of skill in the art, all of which are believed to be within the sphere and scope of the invention as defined by the claims.