Abstract:
In accordance with the present invention, an auto-biased cascode current circuit capable of improved range in headroom is disclosed. In one embodiment, the current circuit includes a current mirror and a bias circuit, where the current mirror contains a reference leg and an output leg. A reference current flows within the reference leg. Included in the output leg is an output terminal, a first output transistor and a second output transistor. The output terminal operates at an output potential. The bias circuit regulates the reference leg of the current mirror such that the output potential is substantially equal to a drain-to-source saturation voltage of the first output transistor plus a drain-to-source saturation voltage of the second output transistor plus a predetermined overdrive voltage. The predetermined overdrive voltage is a design parameter which is less than a threshold voltage. Even as the reference current changes, the bias circuit regulates the reference leg so that the reference current may change significantly while the bias circuit still maintains a proper output potential. In another embodiment, a method for auto-biasing a cascode current circuit is disclosed. The method detects at least one voltage potential from the reference leg and uses this information generate a cascode potential to bias the reference leg.

Description:
FIELD OF THE INVENTION 
     The present invention relates to current mirror circuits, and in particular, to bias circuits for current mirror circuits. 
     BACKGROUND OF THE INVENTION 
     Current circuits of various configurations are a common building block of electronic circuits. Typically, current circuits are used to form a current mirror. Current mirrors either sink or source current in such a way as to respectively receive or provide a substantially constant current to a load. 
     With reference to FIG. 1, a conventional two-transistor current mirror  100  is shown in schematic form. A reference current I refl  is provided to a diode-connected reference transistor MN 1  which is mirrored by an output transistor MN 2  to produce an output current I out1 . Characteristic of current mirrors, the output current I out1  is substantially equal to the reference current I ref1  long as the geometry of the reference transistor MN 1  is substantially the same as the geometry of the output transistor MN 2 . Those skilled in the art can appreciate however, that the ratio of the output current I out1  to the reference current I ref1  may be modified by changing the ratio of the geometry of the output transistor MN 2  to the reference transistor MN 1 . 
     The simple current mirror  100  allows for low-swing operation of an output voltage V out1  of a load, but suffers from poor output resistance. FIG. 2 is a graph which shows the relationship between the output current I out1  along the ordinate direction and the output voltage V out1  along the abscissa. The response graph of the current mirror  100  is divided between a triode region  200  and a saturation region  204 . The saturation region  204  is defined as the output voltage V out1  being larger then a saturation voltage V DS(sat)2  of the output transistor MN 2 . In general, the saturation voltage V DS(sat)  is defined as the drain-to-source voltage of a transistor necessary to begin operation of that transistor in the saturation region which is shown as the “knee” of the curve in FIG.  2 . While operating in the saturation region  204 , changes in output voltage V out1  at the load have little effect on the output current I out1 . However, while operating in the triode region  200 , changes in output voltage V out1 at the load have great effect on the output current I out1 . In other words, the output voltage V out1  can swing as low as the saturation voltage V DS(sat)2  before the output resistance becomes unacceptably affected. Although the simple current mirror  100  provides for a low-swinging output voltage, those skilled in the art can appreciate, that the output resistance is still undesirably low while operating in the saturation region  204 . 
     With reference to FIG. 3, a conventional cascode current mirror  300  is drawn in schematic form. A first reference transistor MN 3  and second reference transistor MN 4 , which are diode connected, form the reference leg  308  of the cascode current mirror while a first output transistor MN 5  and second output transistor MN 6  form the output leg  312 . The second output transistor MN 6  is known as a cascode transistor and serves to buffer output voltage V out2  swings from the first output transistor MN 5  such that the first output transistor MN 5  is more likely to remain operating in saturation. 
     Conventional cascode current mirrors  300  provide excellent output resistance at the expense of a lower swing on the output voltage V out2  (i.e., the ability of the output voltage V out2  to swing low while maintaining a high output resistance). With reference to FIG. 4, a graph of the relationship between output current I out2  along the ordinate direction and output voltage V out2  along the abscissa is shown. When both the first and second output transistors MN 5 , MN 6  are in the saturation region  408 , the output current I out2  remains nearly constant as the output voltage V out2  changes. In other words, the output resistance is extremely high while the output transistors MN 5 , MN 6  are saturated. However, as the second output transistor MN 6  passes into the triode region  404  the output resistance decreases. The output resistance decreases further when both the first and second output transistors MN 5 , MN 6  pass into the triode region  400 . For both output transistors MN 5 , MN 6  to remain in saturation  408 , Equation 1 must be satisfied: 
     
       
         V out(min)2 &gt;V t +V DS(sat)5 +V DS(sat)6   (1) 
       
     
     Equation 1 merely states the minimum output voltage V out(min)2  cannot fall below the sum of a threshold voltage V t , the saturation voltage V DS(sat)5  of the first output transistor MN 5  and the saturation voltage V DS(sat)6  of the second output transistor MN 6 . Where the voltage threshold term V t  is a process variable which is generally the same for all NMOS transistors for a particular semiconductor process and can be defined by the following Equation 2: 
     
       
         V t =V GS −V DS(sat)   (2) 
       
     
     Where V GS  is the gate-to-source voltage of a transistor. Stated another way, the threshold voltage V t  defines the gate-to-source voltage V GS  at which a conduction channel forms between the drain and source. If however, the output voltage falls below the point defined by Equation 1, at least one of the output transistors MN 5 , MN 6  will begin operating in the triode region which significantly decreases the output resistance. It should be noted, that although the output resistance of the cascode current mirror  300  is greater than that of the simple current mirror  100 , the low-swing of the cascode current mirror  300  is considerably higher than the low-swing of the simple current mirror  100 . 
     Output resistance of a current mirror is important because it defines how the output current will change as the output voltage changes. Operating the transistors of the output leg MN 2 , MN 5 , MN 6  of a current mirror  100 ,  300  in the saturation region significantly increases the output resistance. Additionally, the use of the cascode current mirror  300  increases the output resistance when compared to the simple current mirror  100 . 
     Headroom is important because it defines the range in which the output voltage V out2  may operate. The lowest swing of the output voltage V out(min)2  defines the lower limit of the headroom, while the positive power supply V DD  generally defines the upper limit of the headroom (i.e., V out(max)2 =V DD ). Any load circuit which uses the current mirror generally operates within the range defined by the headroom to assure adequate output resistance. Recently, there has been a trend toward lower voltage power supplies V DD , because of their reduced power consumption. However, reducing the power supply V DD  impinges upon the upper range of the headroom V out(max)2  available to the load circuit utilizing the current mirror. Accordingly, there is a need to increase headroom for current mirrors without reducing output resistance. 
     SUMMARY OF THE INVENTION 
     In accordance with the present invention, an auto-biased cascode current circuit capable of improved range in headroom is disclosed. In one embodiment, the current circuit includes a current mirror and a bias circuit, where the current mirror contains a reference leg and an output leg. A reference current flows within the reference leg. Included in the output leg is an output terminal, a first output transistor and a second output transistor. The output terminal operates at an output potential. The bias circuit regulates the reference leg of the current mirror such that the output potential is substantially equal to a drain-to-source saturation voltage of the first output transistor plus a drain-to-source saturation voltage of the second output transistor plus a predetermined overdrive voltage. The predetermined overdrive voltage is a design parameter which is less than a threshold voltage. Even as the reference current changes, the bias circuit regulates the reference leg so that the reference current may change significantly while the bias circuit still maintains a proper output potential. 
     In another embodiment, a method for auto-biasing a cascode current circuit is disclosed. The method detects at least one voltage potential from the reference leg and uses this information to generate a cascode potential to bias the reference leg. In this way, low-swing operation of the cascode current circuit is maintained even if the reference current changes. 
     Based upon the foregoing summary, a number of important advantages of the present invention are readily discerned. A high output resistance is achieved because of the cascode configuration of the current mirror while still allowing the output voltage to swing low. The ability to swing low provides additional range in headroom for the load. Additionally, the current mirror is auto-biased such that a large range of reference currents are supported without needing to redesign the bias circuitry. 
     Additional advantages of the present invention will become readily apparent from the following discussion, particularly when taken together with the accompanying drawings. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a schematic of a conventional current mirror configured to sink current; 
     FIG. 2 is a graph depicting the output current as the output voltage changes for the conventional current mirror of FIG. 1; 
     FIG. 3 is a schematic of a conventional cascode current mirror configured to sink current; 
     FIG. 4 is a graph depicting the output current as the ouput voltage changes for the conventional cascode current mirror of FIG. 3; 
     FIG. 5 is a schematic of a manually-biased cascade current mirror which is capable of supporting a low swinging output voltage; 
     FIG. 6 is a graph depicting the output current as the output voltage changes for the low-swing cascode current mirror of FIG. 5; 
     FIG. 7 is a schematic of a cascode current mirror which features a low swing and auto-biasing; 
     FIG. 8 is a schematic an amplifier circuit which is auto-biased; and 
     FIG. 9 is a schematic representation of a feedback loop equivalent to the auto-biasing circuit of FIG.  7 . 
    
    
     DETAILED DESCRIPTION 
     With reference to FIG. 5, a cascode current mirror  500  with a high output resistance and a low swing output voltage is shown in schematic form. The first and second reference transistors MN 7 , MN 8 , which form a reference leg  508 , are configured such that the output voltage V out(min)3  can swing lower than a conventional cascode current mirror  300  (see FIG.  3 ). More specifically, so long as the minimum output voltage V out(min)3  is such that Equation 3 is satisfied, a first and second output transistors MN 9 , MN 10  of a output leg  512  will remain in saturation V DS(sat)9 , V DS(sat)10 . 
     
       
         V out(min)3 &gt;V DS(sat)9 +V DS(sat)10   (3) 
       
     
     By operating the first and second output transistors MN 9 , MN 10  in the saturation region, the output resistance advantageously remains large. Comparison of Equation 3 with Equation 1, which define the minimum output voltage V out(min)  for their respective circuits, reveals the low swing current mirror  500  can tolerate a lower output voltage V out(min)  than the convention current mirror  300  by an additional voltage threshold V t  while maintaining the same large output resistance. By lowering the swing of the output voltage V out(min)3  for the low-swing current mirror  500 , the range of headroom available to the load is increased accordingly. 
     FIG. 6 shows a graph of an output current I out3  in the ordinate direction and the output voltage V out3  along the abscissa for the low-swing cascode current source  500 . As can be seen from the graph, the output current I out3  remains substantially constant as the output voltage V out3  varies, so long as a first output transistor MN 9  and a second output transistor MN 10  both operate in saturation mode  608 . That as to say, operating the transistors MN 9 , MN 10  in the output leg  512  of the current mirror advantageously provides a large output resistance while both transistors operate in saturation mode  608 . The output resistance decreases when either one  604  or both  600  of the output transistors MN 9 , MN 10  operate in the triode region. 
     Although providing lower swing on the output voltage V out(min)3  and a large output resistance, the cascode current mirror  500  shown in FIG. 5 requires a manual bias circuit  504  to provide a cascode voltage V cas1  to the gate terminal of each of the cascode transistors MN 8 , MN 10 . The optimal minimum value for the cascode voltage V cas1(min)  (i.e., producing the most headroom for the output voltage V out3 ) is the saturation voltage V DS(sat)7  for the first reference transistor MN 7  plus the saturation voltage V DS(sat)8  for the second reference transistor MN 8  plus the threshold voltage V t  for the second reference transistor MN 8 , as defined by the following Equation 4: 
     
       
         V cas1(min) =V DS(sat)7 +V DS(sat)8 +V t   (4) 
       
     
     To produce the cascode voltage V cas1 , a bias current I bias  is provided to a diode connected transistor MN 11  so that the cascode voltage V cas1  properly biases the cascode transistors MN 8 , MN 10 . The bias current I bias  flowing through the diode connected transistor MN 11  forces a proportional gate potential V G11  which is used as the cascode voltage V cas1 . Biasing in this way, allows achieving the low swing of the output voltage V out(min)3  defined by Equation 3 which maximizes the headroom available to the load. 
     To provide a proper bias current I bias  a designer must provide a current source circuit. Generally, these circuits are static. This means they provide a single bias current I bias  which cannot respond to changing needs of the cascode voltage V cas1 . As those skilled in the art can appreciate however, if the reference current I ref3  changes, the saturation voltage V DS(sat)7  must also change to maintain maximum headroom for the output voltage V out3 . As shown in Equation 4 above, the cascode voltage V cas1  should be adjusted when the saturation voltage V DS(sat)7  changes which also means the current source circuit providing the bias current I bias  should change accordingly. It should be noted however, that some applications require accommodation of especially large current swings on the output leg  512  of tune current mirror (i.e., large swings in output current I out3 ) such as switching loads. Large variances in output current I out3  require large swings in reference current I ref3  which require large swings in bias current I bias . 
     As those skilled in the art can appreciate, choosing the proper cascode voltage V cas1  can be an arduous task since the saturation voltage V DS(sat)7  is not only affected by changes in the reference current I ref3  (as discussed above), but also semiconductor process variables, operating temperature, and other factors. Designers typically raise the bias current I bias  to compensate for changes in the reference current I ref3 , semiconductor process variables, operating temperature, and other factors which may affect the saturation voltage V DS(sat)7  and also raise the cascode voltage V cas1 . By raising the cascode voltage V cas1  however, the minimum swing available to the output voltage V out(min)3  also undesirably raises which affects the range of headroom available to the load. This reduction in the headroom is becoming less acceptable as the power supply voltage V DD  is lowered to conserve power. Accordingly, there is a need to provide a low-swing cascode current source which automatically compensates for such factors as the reference current I ref3 , semiconductor process variables and operating temperature. 
     With reference to FIG. 7, an embodiment of an auto-biased low-swing current mirror is shown in schematic form. This embodiment generally includes a cascode current mirror  700  having a reference leg  708  and an output leg  712 , but also includes an auto-biasing circuit  704  which compensates for the factors which require adjusting a cascode voltage Vcas 2  to maintain the maximum range of headroom on the output voltage V out4 . In brief, a first through fourth bias transistors MN 16 , MP 1 , MP 2 , MN 17  of the auto-biasing circuit  704  cooperate to provide feedback which dynamically compensates for such factors as reference current I ref4 , semiconductor process variables and operating temperature in order to properly bias a current mirror  700  portion of the circuit. Use of feedback in this way generally allows for providing the maximum range of headroom to the output voltage V out4  of the load. 
     The goal of the bias circuit  704  is to maintain a minimum headroom voltage V out(min)4 , while factors which affect a saturation voltage V DS(sat)14 , V DS(sat)15  of a first output transistor MN 14  and a second output transistor MN 15  change. The minimum output voltage V out(min)4  which assures the first and second output transistors MN 14  MN 15  remain in saturation V DS(sat)14 , V DS(sat)15  is described in Equation 5: 
     
       
         V out(min)4 &gt;V DS(sat)14 +V DS(sat)15   (5) 
       
     
     As described more fully above, keeping the first and second output transistors MN 14 , MN 15  in saturation desirably creates a large output resistance for the load. 
     To maintain the condition defined in Equation 5 while the factors which affect the saturation voltages V DS(sat)14 , V DS(sat)15  change, a cascode voltage V cas2  and a bias voltage V bias  must also change. If the following Equations 6, 7 and 8 are satisfied, the minimum output voltage defined by Equation 5 is generally maintained: 
     
       
         V bias =V t +V DS(sat)12   (6) 
       
     
     
       
         V cas2(min) =V DS(sat)12 +V DS(sat)13 +V t   (7) 
       
     
     
       
         V D12 =V DS(sat)12   (8) 
       
     
     Where V DS(sat)12  is the saturation voltage of a first reference transistor MN 12  for particular reference current I ref4 , and V D12  is the voltage on the drain of MN 12 . The bias circuit  704  generally satisfies the conditions expressed in Equations 6, 7 and 8 while allowing the reference current I ref4  to preferably change by orders of magnitude. As can be appreciated by those skilled in the art, the auto biasing circuit  704  avoids having to redesign the current source needed to supply a bias current I bias  to the manual bias circuit  504  (see FIG. 5) to accommodate different reference currents I ref3 . 
     The auto bias circuit  704  is comprised of a first through fourth bias transistors MN 16 , MP 1 , MP 2 , MN 17 . The gate of a first bias transistor MN 16  is attached to the drain of the second reference transistor MN 13  and to the gate of the first reference transistor MN 12 . The source of the first bias transistor MN 16  is attached to the source of the second reference transistor MN 13  and to the drain of the first reference transistor MN 12 . A NMOS transistor threshold V t  is produced across the gate and source of the first bias transistor MN 16  (i.e., V GS =V t ). Consequently, the interconnections between the first bias transistor MN 16  and the first and second reference transistors, MN 13  assure a positive transistor threshold +V t  will also exist across the drain and source of the second reference transistor (i.e., V DS13 =V t ), while a negative transistor threshold −V t  wilt exist across the gate and drain of the first transistor (i.e., V GD12=−V   t ). The first bias transistor is matched to the first reference transistor MN 12  (i.e., has substantially the same layout and geometry). 
     The second and third bias transistors MP 1 , MP 2 , are PMOS transistors which form a simple current mirror to source current. The second bias transistor MP 1  is diode connected. Because of the nature of the current mirror, the current through the first bias transistor MN 16  is substantially equal too the current through a fourth bias transistor MN 17 . 
     The fourth bias transistor MN 17  is diode connected. A cascode voltage V cas2  is produced at the gate of the fourth bias transistor MN 17  which is proportional to the current flowing through the fourth bias transistor MN 17 . The cascode voltage V cas2  is provided to the gates of the second reference transistor MN 13  and the second output transistor MN 15 . In this way, the current which flows through the first bias transistor MN 16  affects the cascode voltage V cas2 . 
     The bias circuit  704  uses feedback sensed by the first bias transistor MN 16  to set the cascode voltage V cas2 . There are two modes of operation for the bias circuit  704  in which the loop gain of the feedback loop is different. When the drain-to-source voltage V DS13  of the second reference transistor MN 13  is less that the voltage threshold V t , the first bias transistor MN 16  allows less current to flow, limits the feedback and decreases the cascode voltage V cas2 . Alternatively, when the drain-to-source voltage V DS13  of the second reference transistor MN 13  is more that the voltage threshold V t , the first bias transistor MN 16  allows more current to flow, increases the feedback and increases the cascode voltage V cas2 . The cascode voltage V cas2  applied to the second reference transistor MN 13  affects the drain-to-source voltage V DS13  of the second reference transistor MN 13  such that the feedback loop as complete. 
     As those skilled in the art can appreciate, a current mirror may be configured as a voltage amplifier. With reference to FIG. 8, an embodiment of a voltage amplifier leg  800  which utilizes the present invention is shown. Changes on the input voltage V in  are reflected in the output voltage V out5  and output current I out5  such that the amplifier leg  800  is characterized as having a gain. It should be noted, the same reference  708  and bias circuitry  704  are used to properly bias he amplifier leg  800 . The ability to auto-bias this amplifier allows low-swing operation of the amplifier leg  800 . 
     With reference to FIG. 9, the bias circuit  704  is represented as block diagram of a feedback loop. The feedback loop receives the drain-to-source voltage V DS13  of the second reference transistor MN 13  as an input  900  to produce the cascode voltage V cas2  as an output  904 . A dual mode gain block  908  is applied to the input  904 . As explained above, the value of the drain-to-source voltage V DS13  of the second reference transistor MN 13  dictates whether the first bias transistor MN 16  passes a large current or a small current which is represented as the dual mode gain block  908 . A feedback block  912  reflects changes in the cascode voltage V cas2  as changes in the drain-to-source voltage V DS13  of the second reference transistor MN 13 . As can be appreciated by those skilled in the art, changes in the gate-to-source potential of a transistor will cause changes in the drain-to-source voltage. In this way, the output of the feedback loop  904  settles into supplying the saturation voltage V DS(sat)12  of the first reference transistor MN 13  to the gate of the second reference transistor MN 13  even if the reference current I ref4  changes the saturation voltage V DS(sat)12 . 
     Often designers wish to provide excess bias to the drain of the first reference transistor MN 12 . This concept is sometime referred to by those skilled in the art as saturation voltage overdrive V overdrive . When a transistor is biased at the “knee” of the saturation region it is said to be at the saturation voltage V DS(sat) , however, applying an extra amount of bias to the drain (i.e., applying voltage overdrive V overdrive ) will insure that the transistor is biased beyond the “knee” and will likely remain in the saturation region. Reference current I ref4  changes, semiconductor process variances, operating temperature changes, and other factors can be additionally compensated for by providing for saturation voltage overdrive V overdrive . 
     The bias circuit  704  is capable of providing extra bias V overdrive  to the cascode voltage V cas2  such that the first reference transistor MN 12  is more likely to remain in saturation as conditions change. Providing saturation voltage overdrive V overdrive  is accomplished by making the fourth bias transistor MN 17  weak with respect to the first bias transistor MN 16 . Since the current flowing in each leg of the current source of the bias circuit  704  is generally equal because of the current mirror defined by the second and third bias transistors MP 1 , MP 2 , the gate voltage V G17  of the fourth bias transistor MN 17  must increase to accommodate the current, if the device is made weaker. By increasing the gate voltage V G17 , the cascode voltage V cas2  also increases which provides saturation voltage overdrive V overdrive  to the first reference transistor MN 12 . 
     Although the above discussion is generally limited to current mirrors configured as current sinks, those skilled in the art can appreciate the principals are equally applicable to current sources as well. Additionally, while the embodiments disclosed use CMOS transistors, the concepts are equally applicable to other transistor types. 
     The forgoing description of the invention has been presented for the purposes of illustration and description and is not intended to limit the invention. Variations and modifications commensurate with the above description, together with the skill or knowledge of the relevant art, are within the scope of the present invention. The embodiments described herein are further intended to explain the best mode known for practicing the invention and to enable those skilled in the art to utilize the invention in such best mode or other embodiments, with the various modifications that may be required by the particular application or use of the invention. It is intended that the appended claims be construed to include alternative embodiments to the extent permitted by the prior art.