Abstract:
An Orthogonal Frequency Division Multiplexing (OFDM) communiation system and method for improving frequency utilization efficiency. In the system, a Reed-Solomon encoder codes input information data, and ouputs a Reed-Solomon block having a second number of Reed-Solomon symbols, each Reed-Solomon symbol having a first number of Reed-Solomon symbol elements. An interleaver receives the Reed-Solomon block, and disperses the Reed-Solomon symbol elements existing in a specified one Reed-Solomon symbol within the received Reed-Solomon block in the same sub-channel position in a fourth number of sub-channels of each of a third number of consecutive OFDM symbols.

Description:
PRIORITY 
     This application claims priority to an application entitled “OFDM Communication System and Method for Improving Data Transmission Performance” filed in the Korean Industrial Property Office on Mar. 27, 2001 and assigned Ser. No. 2001-16019, the contents of which are hereby incorporated by reference. 
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates generally to an OFDM (Orthogonal Frequency Division Multiplexing) scheme, and in particular, to an OFDM communication system and method for improving frequency utilization efficiency. 
     2. Description of the Related Art 
     An OFDM scheme recently used for high-speed data transmission over wired/wireless channels transmits data using multiple carriers. The OFDM scheme is a kind of an MCM (Multi-Carrier Modulation) scheme, which converts a serial input symbol stream to parallel symbol streams, and modulates the symbol streams with a plurality of orthogonal sub-carriers (or sub-channels) before transmission. 
     The MCM scheme was first applied to an HF (High Frequency) radio for military use in the late 1950&#39;s, and the OFDM scheme overlapping a plurality of orthogonal sub-carriers has been developed from 1970&#39;s. Since it is difficult to implement orthogonal modulation between multiple carriers, the application of the MCM and OFDM schemes to an actual system is limited. However, since Weinstein et al. announced in 1971 that OFDM modulation/demodulation could be efficiently processed using DFT (Discrete Fourier Transform), the technical development of the OFDM scheme has made rapid progress. In addition, as the use of a guard interval and a method of inserting a cyclic prefix guard interval are known, the negative effects of the system on multiple paths and delay spread have decreased further. Therefore, the OFDM scheme is widely applied to the digital transmission technologies such as digital audio broadcasting (DAB), digital television, wireless local area network (WLAN), and wireless asynchronous transfer mode (WATM). That is, although the OFDM scheme was not widely used due to its hardware complexity, recent development of various digital signal processing technologies including fast Fourier transform (FFT) and inverse fast Fourier transform (IFFT) makes it possible to implement the OFDM scheme. Though similar to the conventional FDM (Frequency Division Multiplexing) scheme, the OFDM scheme is characterized in that it can obtain optimal transmission efficiency during high-speed data transmission by maintaining orthogonality among a plurality of sub-carriers. In addition, the OFDM scheme has excellent frequency efficiency and is resistant to multi-path fading, thus making it possible to obtain optimal transmission efficiency during high-speed data transmission. Further, since the OFDM scheme uses overlapped frequency spectrums, it has excellent frequency utilization efficiency, is resistant to frequency selective fading, is resistant to multi-path fading, can reduce the effects of ISI (Inter-Symbol Interference) using the guard interval, can simply design the hardware structure of an equalizer, and is resistant to impulse noses. Hence, the OFDM scheme tends to be actively applied to the communication system. 
     Now, a structure of a common OFDM system will be described with reference to  FIG. 1 . 
       FIG. 1  illustrates a structure of an OFDM system according to the prior art. Referring to  FIG. 1 , received information data  101  is provided to an error correction encoder  102 . The error correction encoder  102  codes the received information data  101  using error correction coding previously set in the OFDM system, i.e., Reed-Solomon coding, and provides its output to an interleaver  103 . The interleaver  103  interleaves the output signal of the encoder  102  for preventing burst errors, and provides its output to a serial-to-parallel (S/P) converter  104 . The S/P converter  104  forms a plurality of sub-channels by arranging serial data output from the interleaver  103  in the form of parallel data, and provides the sub-channels to a pilot adder  106 . The pilot adder  106 , under the control of a pilot controller  105 , adds pilots to the sub-channels output from the S/P converter  104 , and provides the pilot-added sub-channels to a sub-channel mapper  107 . Here, the pilot controller  105  generates pilot data blocks by phase-shifting a plurality of pilot data blocks previously set in the OFDM system with a random code. The pilot adder  106  adds the pilot data blocks generated by the pilot controller  105  to the pilot sub-channels, and outputs K sub-channels [C( 1 ), C( 2 ), . . . , C(K)] along with a plurality of sub-channels. 
     The sub-channel mapper  107  performs signal-mapping on constellation for the K sub-channels output from the pilot adder  106 , and outputs signal-mapped sub-channels [S( 1 ), S( 2 ), . . . , S(K)]. Here, the signal mapping may be performed according to BPSK (Binary Phase Shift Keying), QPSK (Quadrature Phase Shift Keying), 16QAM (16-ary Quadrature Amplitude Modulation) or 64QAM modulation. The signal-mapped signals [S( 1 ), S( 2 ), . . . , S(K)] output from the sub-channel mapper  107  are provided to an inverse fast Fourier transformer (IFFT)  108 . Here, the IFFT  108 , a K-point inverse fast Fourier transformer, OFDM-multiplexes the signals output from the sub-channel mapper  107  and provides the OFDM-multiplexed signals [s( 1 ), s( 2 ), . . . , s(K)] to a parallel-to-serial (P/S) converter  109 . The P/S converter  109  converts the OFDM-multiplexed signals [s( 1 ), s( 2 ), . . . , s(K)] in the form of parallel data output from the IFFT  108  into a serial signal, and outputs the serial signal as output data  110 . 
     Compared with other systems, the OFDM system having the structure described in conjunction with  FIG. 1  has excellent frequency utilization efficiency and is resistant to multi-path fading and frequency selective fading. However, there is a need for an OFDM system having more excellent frequency utilization efficiency and is more resistant to the multi-path fading and frequency selective fading. 
     SUMMARY OF THE INVENTION 
     It is, therefore, an object of the present invention to provide an interleaving apparatus and method for improving transmission error performance on Reed-Solomon coded symbols. 
     It is another object of the present invention to provide a sub-channel repetition apparatus and method for improving transmission error performance by repeatedly transmitting the same data over a plurality of different sub-channels. 
     It is further another object of the present invention to provide a sub-channel repetition apparatus and method for removing frequency selective fading. 
     It is yet another object of the present invention to provide a sub-channel assignment apparatus and method for acquiring frequency diversity using sub-channel frequency transition. 
     It is still another object of the present invention to provide an apparatus and method for transmitting sub-channels having a minimized PAPR (Peak-to-Average Power Ratio). 
     It is still another object of the present invention to provide an apparatus and method for detecting transmitted sub-channels having a minimized PAPR without using separate supplemental information. 
     It is still another object of the present invention to provide a system and method for acquiring antenna diversity. 
     In accordance with a first aspect of the present invention, there is provided a system for improving error correction capability in an OFDM (Orthogonal Frequency Division Multiplexing) communication system. The system comprises a Reed-Solomon encoder for coding input information data, and outputting a Reed-Solomon block comprised of a second number of Reed-Solomon symbols each comprised of a first number of Reed-Solomon symbol elements; and an interleaver for receiving the Reed-Solomon block, and dispersing the Reed-Solomon symbol elements existing in a specified one Reed-Solomon symbol within the received Reed-Solomon block in the same sub-channel positions in a fourth number of sub-channels of each of a third number of consecutive OFDM symbols. 
     In accordance with a second aspect of the present invention, there is provided a system for repeatedly transmitting sub-channels in an OFDM communication system. The system comprises a sub-channel repeater for repeating input data blocks so as to transmit each of the input data blocks over a predetermined number of sub-channels; and a plurality of mappers for mapping the sub-channels output from the sub-channel repeater according to a predetermined modulation mode. 
     In accordance with a third aspect of the present invention, there is provided a system for performing sub-channel assignment in an OFDM communication system. The system comprises a plurality of selectors for selecting a specific sub-channel data block among input sub-channel data blocks according to a control signal, and transmitting the selected sub-channel data block over a corresponding sub-channel; and a sub-channel assignment controller for controlling sub-channel assignment such that each of the selectors converts a sub-channel data block to be selected from the sub-channel data blocks in a predetermined period of time. 
     In accordance with a fourth aspect of the present invention, there is provided a system for transmitting sub-channels having a minimum PAPR (Peak-to-Average Power Ratio) in on OFDM communication system. The system comprises a pilot scrambling code generator for generating a predetermined number of pilot scrambling codes for identifying pilot sub-channel data blocks among input sub-channel data blocks; a scrambling code generator for generating a predetermined number of scrambling codes for scrambling the input sub-channel data blocks; a plurality of first multipliers for multiplying the input pilot sub-channel data blocks by a first pilot scrambling code among the pilot scrambling codes, for scrambling; a plurality of second multipliers for multiplying the sub-channel data blocks excluding the pilot sub-channel data blocks from the input sub-channel data blocks and data blocks output from the first multipliers by a first scrambling code among the scrambling codes, for scrambling; a first inverse fast Fourier transformer (IFFT) for IFFT-transforming the signals output from the second multipliers; a plurality of third multipliers for multiplying the input pilot sub-channel data blocks by a second pilot scrambling code among the pilot scrambling codes, for scrambling; a plurality of fourth multipliers for multiplying the sub-channel data blocks excluding the pilot sub-channel data blocks from the input sub-channel data blocks and data blocks output from the third multipliers by a second scrambling code among the scrambling codes, for scrambling; a second IFFT for IFFT-transforming the signals output from the fourth multipliers; first and second PAPR calculators for calculating PAPRs of the sub-channel data blocks output from the first IFFT and the second IFFT, respectively; and a selector for selecting sub-channel data blocks output from the first and second IFFTs having a minimum PAPR among the calculated PAPRS, and transmitting the selected sub-channel data blocks over a sub-channel of the OFDM communication system. 
     In accordance with a fifth aspect of the present invention, there is provided a transmission system employing transmission antenna diversity in an OFDM communication system. The system comprises a first antenna for transmitting an in-phase signal having no phase offset with output data, upon receiving the output data; and a second antenna for alternately transmitting the received output data as an in-phase signal having no phase offset with the output data and as a phase-inversed signal having a 180°-phase offset with the output data in a training symbol period. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The above and other objects, features and advantages of the present invention will become more apparent from the following detailed description when taken in conjunction with the accompanying drawings in which: 
         FIG. 1  illustrates a structure of an OFDM system according to the prior art; 
         FIG. 2  illustrates a structure of an OFDM system performing a function according to an embodiment of the present invention; 
         FIG. 3  illustrates a structure of Reed-Solomon coded data symbols according to an embodiment of the present invention; 
         FIG. 4  illustrates an interleaver structure for interleaving Reed-Solomon coded OFDM symbols according to an embodiment of the present invention; 
         FIG. 5  illustrates an OFDM symbol structure and sub-channel arrangement based on BPSK modulation according to an embodiment of the present invention; 
         FIG. 6  illustrates an OFDM symbol structure and sub-channel arrangement based on QPSK modulation according to an embodiment of the present invention; 
         FIG. 7  illustrates an OFDM symbol structure and sub-channel arrangement based on 16QAM modulation according to an embodiment of the present invention; 
         FIG. 8  illustrates an OFDM symbol structure and sub-channel arrangement based on 64QAM modulation according to an embodiment of the present invention; 
         FIG. 9  illustrates a structure of a sub-channel repeater according to a first embodiment of the present invention; 
         FIG. 10  illustrates a structure of a sub-channel repeater according to a second embodiment of the present invention; 
         FIG. 11  illustrates an internal structure of the sub-channel repeater shown in  FIGS. 9 and 10 ; 
         FIGS. 12A and 12B  illustrate a sub-channel assignor according to an embodiment of the present invention; 
         FIG. 13  illustrates an internal structure of a sub-channel assignor according to an embodiment of the present invention; 
         FIG. 14  illustrates a structure of a minimum PAPR select sub-channel transmitter according to an embodiment of the present invention; 
         FIG. 15  illustrates a structure of an extended minimum PAPR select sub-channel transmitter in which the number of IFFTs is extended; 
         FIG. 16  illustrates a structure of a receiver corresponding to the minimum PAPR select sub-channel transmitter of  FIG. 15 ; and 
         FIG. 17  illustrates a transmission diversity scheme according to an embodiment of the present invention. 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     A preferred embodiment of the present invention will be described herein below with reference to the accompanying drawings. In the following description, well-known functions or constructions are not described in detail since they would obscure the invention in unnecessary detail. 
     The present invention provides five embodiments for improving OFDM communication performance, i.e., the frequency utilization efficiency and the multi-path fading characteristic. A brief description of the five embodiments will be given below. 
     (1) First Embodiment 
     The first embodiment proposes an interleaving apparatus and method for improving system performance by improving error correction performance of the OFDM system, when the OFDM system codes transmission information data by Reed-Solomon coding. The first embodiment interleaves/deinterleaves data symbols such that a group of error (or damaged) data blocks is arranged in a specified one of Reed-Solomon coded symbols. That is, this embodiment improves error correction capability for the frequency selective fading by performing interleaving and deinterleaving such that respective data blocks in one Reed-Solomon symbol should be mapped to the same sub-channels in a plurality of OFDM symbols. 
     (2) Second Embodiment 
     The second embodiment provides an apparatus and method for performing repetitive transmission on a plurality of different OFDM sub-channels in the OFDM system. By performing repetitive transmission on the sub-channels, it is possible to acquire frequency diversity. Hence, the OFDM system provides reliable data communication even in a frequency selective fading environment or a poor environment where an intended/non-intended interference signals exist. Further, it is possible to perform channel mapping such that during repetitive transmission, the associated sub-channels vary depending on the time, thus acquiring additional frequency diversity. 
     (3) Third Embodiment 
     The third embodiment provides a scheme for dynamically performing OFDM sub-channel assignment according to a predetermined code pattern or a pattern previously set in the OFDM system depending on the time, rather than statically performing sub-channel mapping, or adaptively performing the sub-channel assignment according to the channel condition. Since the sub-channel frequency is not static but dynamic, it is possible to acquire frequency diversity. 
     (4) Fourth Embodiment 
     The fourth embodiment provides a method for detecting a selected sub-channel with the minimized PAPR (Peak-to-Average Power Ratio) using a plurality of scrambling codes at a receiver, without transmitting separate supplemental information at a transmitter in the OFDM system. The minimization of the PAPR reduces a load of a power amplifier (PA) in the transmitter, making it possible to readily implement the power amplifier. In addition, the method according to the fourth embodiment of the present invention performs IFFT (Inverse Fast Fourier Transform) by scrambling transmission data using a plurality of predetermined codes (complementary codes in this embodiment) by the transmitter, and selecting the sub-channel having the minimum PAPR. In the prior art scheme, the transmitter transmits transmission data along with supplemental information for the scrambling code having minimum PAPR, so that the receiver detects the supplemental information. However, in the embodiment of the present invention, even though the transmitter does not transmit the supplemental information for the scrambling code, the receiver can detect the sub-channel selected by the transmitter, thus contributing to simplification of the hardware structure of the transceiver. Further, since it is not necessary to transmit the supplemental information, additional overhead is not required. 
     (5) Fifth Embodiment 
     The fifth embodiment provides a scheme for implementing transmission antenna diversity for alternating phases in a training symbol period so that the receiver can estimate the characteristics of different transmission channels when diversity is applied to the transmission antennas. In the fifth embodiment of the present invention, the OFDM system having a plurality of transmission antennas, e.g., 2 transmission antennas, transmits an in-phase signal (of 0 degree phase) with a first antenna, and alternately transmits signals with a second antenna in the training symbol period. That is, the OFDM system first transmits an in-phase signal (of 0 degree phase) and next transmits a phase-inversed signal (of 180 degree phase). Accordingly, the receiver can perform channel estimation on the respective transmission paths used by the transmitter in transmitting the signals through the two antennas, and performs data processing and demodulation using the estimation results on the respective transmission channels, thus improving system performance. 
     A detailed description of the first to fifth embodiments will be made with reference to the accompanying drawings. 
       FIG. 2  illustrates a structure of an OFDM system performing a function according to an embodiment of the present invention. Referring to  FIG. 2 , input transmission information data  201  is switched to a convolutional encoder  202  according to a control signal. The convolutional encoder  202  convolutional-codes the input information data  201 , and provides its output to an interleaver  203 . The interleaver  203  interleaves the signal output from the convolutional encoder  202  according to a preset interleaving rule, and provides its output to an S/P converter  206 . Of course, although the input information data  201  can be subject to convolutional coding, the embodiment of the present invention will be described on the assumption that the input information data  201  is subject to Reed-Solomon coding. Then, the input information data  201  is provided to a Reed-Solomon (RS) encoder  204  under the control of the OFDM system. The Reed-Solomon encoder  204  performs Reed-Solomon coding on the input information data  201 , and provides its output to an interleaver  205 . The interleaver  205  interleaves the signal output from the Reed-Solomon encoder  204  using an interleaving rule based on the first embodiment of the present invention, and provides its output to the S/P converter  206 . The interleaving rule based on the first embodiment, especially an interleaving rule for the Reed-Solomon coded data symbols will be described later with reference to  FIGS. 3 to 8 . 
     The S/P converter  206  converts the interleaved signal in the form of a serial signal into M parallel signals, i.e., parallel signals comprised of a plurality of sub-channels, and provides its output to a sub-channel repeater  207 . A sub-channel repetition operation of the sub-channel repeater  207  is controlled by a repetition controller  208 , and the repetition controller  208  controls the repetitive transmission using channel information  209 . A detailed description of the sub-channel repeater  207  and the repetition controller  208  according to the embodiment of the present invention will be given later with reference to  FIGS. 9 to 11 . 
     The sub-channel repeated signals are provided to a pilot adder  210 . The pilot adder  210 , under the control of a pilot controller  211 , adds pilot sub-channels to the signals output from the sub-channel repeater  207 , and provides its output to a sub-channel assignor  212 . The sub-channel assignor  212 , under the control of a sub-channel assignment controller  213 , receives the signals output from the pilot adder  210  and dynamically adaptively assigns the OFDM sub-channels by varying the sub-channels according to the set time or the service type, rather than statically assigning the sub-channels. The sub-channel assignment controller  213  controls the dynamic/adaptive sub-channel assignment according to the channel condition, using the channel information  209 . A detailed description of the sub-channel assignor  212  and the sub-channel assignment controller  213  will be made later with reference to  FIGS. 12A to 13 . 
     The sub-channel signals output from the sub-channel assignor  212  are provided to a sub-channel mapper  214 . The sub-channel mapper  214 , under the control of a mapping controller  215 , performs mapping for modulation of the respective sub-channels according to a modulation mode determined based on a data rate, and provides the mapped signals to a sub-channel scrambler  216 . Here, the signal mapping may be performed according to BPSK, QPSK, 16QAM or 64QAM modulation. The sub-channel scrambler  216  scrambles the signals output from the sub-channel mapper  214  with a scrambling code generated by a scrambling code controller  217 , and provides the scrambled signals to an inverse fast Fourier transformer (IFFT)  218 . Here, the sub-channel scrambler  216  and the scrambling code controller  217  scramble each OFDM symbol data block by several scrambling codes, rather than simply scrambling the sub-channels, and then provide the scrambled data blocks to the IFFT  218 . Although the OFDM system of  FIG. 2  includes a single IFFT  218 , the OFDM system may include as many IFFTs as the number of scrambling codes, when a plurality of scrambling codes are used. Such a structure will be described later with reference to  FIGS. 14 and 15 . The IFFT-transformed sub-channels are provided to a PAPR calculator &amp; minimum PAPR sub-channel selector  219 . The PAPR calculator &amp; minimum PAPR sub-channel selector  219  receives the signals output from the IFFT  218 , calculates PAPRs of the received signals, and selects the IFFT-transformed signal having the minimum PAPR among the IFFT-transformed signals output from the IFFT  218 . The IFFT  218  provides the sub-channels corresponding to the selected IFFT-transformed signal having the minimum PAPR to a P/S converter  220 . That is, the OFDM system scrambles each OFDM symbol data block with different scrambling codes, subjects the scrambled signals to IFFT, and selects a sequence having minimum PAPR from the IFFT-transformed signals. The embodiment of the present invention provides a scrambling scheme for reducing the PAPR. In this scrambling scheme, even though the transmitter does not transmit separate supplemental information on the scrambling code used by the transmitter itself, the receiver can detect the corresponding sequence using only the pilot sub-channel. This scrambling scheme will be described in detail with reference to  FIGS. 14 and 15 . 
     The P/S converter  220  converts the parallel sub-channel signals output from the PAPR calculator &amp; minimum PAPR sub-channel selector  219  into a serial signal X(t). The signal output from the P/S converter  220  is provided to a transmission diversity device  222 . The transmission diversity device  222  performs transmission diversity to transmit the serial signal X(t) through a plurality of antennas, e.g., 2 antennas. When transmitting the transmission signal using the two transmission antennas, the transmission diversity device  222  transmits an in-phase signal (0 degree phase) with a first antenna, and alternately transmits signals with a second antenna in the training symbol period. That is, the transmission diversity device  222  first transmits an in-phase signal (0 degree phase) and next transmits a phase-inversed signal (180 degree phase). As a result, a first transmission diversity signal X 1 (t) is transmitted through a first antenna ANT 1  ( 223 ), while a second transmission diversity signal X 2 (t) is transmitted through a second antenna ANT 2  ( 224 ). A transmission diversity scheme of the transmission diversity device  222  according to the embodiment of the present invention will be described in detail with reference to  FIG. 16 . 
     Next, a detailed description will be made of the embodiments of the present invention in the OFDM system having the structure described in conjunction with  FIG. 2 . 
     First, a description of a scheme for interleaving the Reed-Solomon coded symbol data will be made with reference to  FIGS. 3 to 8 . 
       FIG. 3  illustrates a structure of Reed-Solomon coded data symbols according to an embodiment of the present invention. As described in conjunction with  FIG. 2 , the OFDM system employing Reed-Solomon coding should arrange Reed-Solomon coded symbol elements in the sub-channels located in the same positions of the OFDM symbols, in order to improve error correction capability of the system. By deinterleaving the OFDM symbols interleaved in this manner, the receiver arranges the error (or damaged) data on the transmission channel in one Reed-Solomon symbol of the Reed-Solomon decoder, thus making it possible to improve error correction capability. Particularly, in a frequency selective fading environment, it is possible to further improve the error correction capability. 
     The data symbol structure based on the Reed-Solomon coding, shown in  FIG. 3 , is an output of the Reed-Solomon encoder  204  having GF(2**8), k Reed-Solomon input symbols, n=48 Reed-Solomon output symbols, and error correction capability of t=(n−k)/2. The output of the Reed-Solomon encoder  204  can be defined as 
     
       
         
           
             
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     Each of Reed-Solomon (RS) blocks [B 1 , B 2 , . . . , B 6 ]  311 – 321  is comprised of 48 Reed-Solomon symbols [S 1 , S 2 , . . . , S 48 ], and since GF(2**8) is used, each Reed-Solomon symbol is comprised of 8 Reed-Solomon elements [b 1 , b 2 , . . . , b 8 ]. The output of the Reed-Solomon encoder  204  having this structure is provided to the interleaver  205  illustrated in  FIG. 4 . 
       FIG. 4  illustrates an interleaver structure for interleaving Reed-Solomon coded OFDM symbols according to an embodiment of the present invention. The interleaver  205  receives an output signal  411  of the Reed-Solomon encoder  204 , and interleaves the received signal  411  with an OFDM signal. The received signal  411  is interleaved such that it is mapped to the OFDM sub-channels. By the interleaving operation of the interleaver  205 , the output signal  411  of the Reed-Solomon encoder  204  is converted into OFDM symbol data and then arranged. Several modulation methods for modulating the interleaved signal will be described with reference to  FIGS. 5 to 8 . 
       FIG. 5  illustrates an OFDM symbol structure and sub-channel arrangement based on BPSK modulation according to an embodiment of the present invention,  FIG. 6  illustrates an OFDM symbol structure and sub-channel arrangement based on QPSK modulation according to an embodiment of the present invention,  FIG. 7  illustrates an OFDM symbol structure and sub-channel arrangement based on 16QAM modulation according to an embodiment of the present invention, and  FIG. 8  illustrates an OFDM symbol structure and sub-channel arrangement based on 64QAM modulation according to an embodiment of the present invention. 
     Referring first to  FIG. 5 , each of OFDM symbols [ 01 ,  02 , . . . ,  08 ]  511 – 519  is comprised of 48 sub-channels [C 1 , C 2 , . . . , C 48 ], and since the BPSK modulation is used, each sub-channel receives 1-bit data. As described in  FIG. 3 , one Reed-Solomon block, e.g., the Reed-Solomon block [B 1 ]  311  is comprised of 48×8 bits, and the 48×8 bits are arranged in the OFDM symbols [ 01 ,  02 , . . . ,  08 ]  511 – 519  as shown in  FIG. 5  by the interleaver  205 . The 8 OFDM symbols  511 – 519  are also comprised of 8×48 bits, which are equal to the bit number of one Reed-Solomon block  311 . Now, a description will be made as to how the interleaver  205  arranges the Reed-Solomon block [B 1 ]  311  in the 8 OFDM symbols  511 – 519 . 
     The interleaver  205  arranges  8  Reed-Solomon elements [b 1 , b 2 , . . . , b 8 ] of a first Reed-Solomon symbol S 1  of the Reed-Solomon block [B 1 ]  311  in first sub-channels [ 01 -C 1 ,  02 -C 1 ,  03 -C 1 , . . . ,  08 -C 1 ] of the 8 OFDM symbols [ 01 ,  02 , . . . ,  08 ]  511 – 519 . That is, the interleaver  205  arranges B 1 -S 1 -b 1  in  01 -C 1 , arranges B 1 -S 1 -b 2  in  02 -C 1 , and arranges B 1 -S 1 -b 8  in  08 -C 1 . That is, the interleaver  205  performs interleaving such that the 8 Reed-Solomon elements of the first Reed-Solomon symbol S 1  are arranged in first sub-channels of the 8 OFDM symbols. Upon receiving such interleaved signal transmitted by the transmitter, the receiver performs inverse interleaving, i.e., deinterleaving. The deinterleaving refers to arranging the same sub-channel data blocks of the 8 OFDM symbols in one Reed-Solomon symbol. Therefore, if there exists frequency selective fading or narrow-band jamming signal on the transmission channel, transmission errors occur in a specific sub-channel of the OFDM symbol. The transmission errors occurred in the specific sub-channel are arranged in one Reed-Solomon symbol by deinterleaving, thus contributing to an improvement in error correction capability of the Reed-Solomon coding. Compared to the case where errors are dispersively arranged in a plurality of Reed-Solomon symbols, arranging the errors in one Reed-Solomon symbol extends error correction capability from 4×1 bits up to 4×8 bits, thus improving the system performance. 
     The interleaving based on the BPSK modulation according to the present invention has been described with reference to  FIG. 5 . Next, interleaving based on the QPSK modulation will be described with reference to  FIG. 6 . 
       FIG. 6  illustrates an OFDM symbol structure and sub-channel arrangement based on QPSK modulation according to an embodiment of the present invention. 
     Referring to  FIG. 6 , each of OFDM symbols [ 01 ,  02 , . . . ,  08 ]  611 – 619  is comprised of 48 sub-channels [C 1 , C 2 , . . . , C 48 ], and since the QPSK modulation is used, each sub-channel receives 2-bit data. As described in  FIG. 3 , two Reed-Solomon blocks, e.g., the Reed-Solomon blocks [B 1 ]  311  and [B 2 ]  313  are comprised of 48×8×2 bits. The interleaver  205  arranges the 2 Reed-Solomon blocks [B 1 ]  311  and [B 2 ]  313  in the 8 OFDM symbols [ 01 ,  02 , . . . ,  08 ]  611 – 619  by interleaving. The 8 OFDM symbols  611 – 619  are also comprised of 2×8×48 bits, which are equal to the bit number of two Reed-Solomon blocks [B 1 ]  311  and [B 2 ]  313 . Now, a description will be made as to how the interleaver  205  arranges the Reed-Solomon blocks [B 1 ]  311  and [B 2 ]  313  in the 8 OFDM symbols  611 – 619 . 
     The interleaver  205  arranges 8 Reed-Solomon elements [b 1 , b 2 , . . . , b 8 ] of a first Reed-Solomon symbol S 1  of the first Reed-Solomon block [B 1 ]  311  and 8 Reed-Solomon elements [b 1 , b 2 , . . . , b 8 ] of a first Reed-Solomon symbol S 1  of the second Reed-Solomon block [B 2 ]  313  in first sub-channels [ 01 -C 1 ,  02 -C 1 ,  03 -C 1 , . . . ,  08 -C 1 ] of the 8 OFDM symbols [ 01 ,  02 , . . . ,  08 ]  611 – 619 . That is, the interleaver  205  arranges B 1 -S 1 -b 1  and B 2 -S 1 -b 1  in  01 -C 1 , arranges B 1 -S 1 -b 2  and B 2 -S 1 -b 2  in  02 -C 1 , and arranges B 1 -S 1 -b 8  and B 2 -S 1 -b 8  in  08 -C 1 . That is, the interleaver  205  performs interleaving such that the 2×8 Reed-Solomon elements of the first Reed-Solomon symbols S 1  in the first Reed-Solomon block [B 1 ]  311  and the second Reed-Solomon block [B 2 ]  313  are arranged in first sub-channels of the 8 OFDM symbols. Upon receiving such interleaved signal transmitted by the transmitter, the receiver performs inverse interleaving, i.e., deinterleaving. The deinterleaving means arranging the same sub-channel data blocks of the 8 OFDM symbols in one Reed-Solomon symbol. Therefore, if there exists frequency selective fading or narrow-band jamming signal on the transmission channel, transmission errors occur in a specific sub-channel of the OFDM symbol. The transmission errors occurred in the specific sub-channel are arranged only in a specified one Reed-Solomon symbol by deinterleaving, contributing to an improvement in error correction capability of the Reed-Solomon coding, thereby improving the system performance. 
     The interleaving based on the QPSK modulation according to the present invention has been described with reference to  FIG. 6 . Next, interleaving based on the 16QAM modulation will be described with reference to  FIG. 7 . 
       FIG. 7  illustrates an OFDM symbol structure and sub-channel arrangement based on 16QAM modulation according to an embodiment of the present invention. 
     Referring to  FIG. 7 , each of OFDM symbols [ 01 ,  02 , . . . ,  08 ]  711 – 719  is comprised of 48 sub-channels [C 1 , C 2 , . . . , C 48 ], and since the 16QAM modulation is used, each sub-channel receives 4-bit data. As described in  FIG. 3 , four Reed-Solomon blocks, e.g., the Reed-Solomon blocks [B 1 , B 2 , B 3 , B 4 ]  311 – 317  are comprised of 48×8×4 bits. The interleaver  205  arranges the 4 Reed-Solomon blocks [B 1 , B 2 , B 3 , B 4 ]  311 – 317  in the 8 OFDM symbols [ 01 ,  02 , . . . ,  08 ]  711 – 719  by interleaving. The 8 OFDM symbols  711 – 719  are also comprised of 4×8×48 bits, which are equal to the bit number of 4 Reed-Solomon blocks [B 1 , B 2 , B 3 , B 4 ]  311 – 317 . Now, a description will be made as to how the interleaver  205  arranges the Reed-Solomon blocks [B 1 , B 2 , B 3 , B 4 ]  311 – 317  in the 8 OFDM symbols  711 – 719 . 
     The interleaver  205  arranges 8 Reed-Solomon elements [b 1 , b 2 , . . . , b 8 ] of a first Reed-Solomon symbol S 1  of the first Reed-Solomon block [B 1 ]  311 , 8 Reed-Solomon elements [b 1 , b 2 , . . . , b 8 ] of a first Reed-Solomon symbol S 1  of the second Reed-Solomon block [B 2 ]  313 , 8 Reed-Solomon elements [b 1 , b 2 , . . . , b 8 ] of a first Reed-Solomon symbol S 1  of the third Reed-Solomon block [B 3 ]  315 , and 8 Reed-Solomon elements [b 1 , b 2 , . . . , b 8 ] of a first Reed-Solomon symbol S 1  of the fourth Reed-Solomon block [B 4 ]  317  in first sub-channels [ 01 -C 1 ,  02 -C 1 ,  03 -C 1 , . . . ,  08 -C 1 ] of the 8 OFDM symbols [ 01 ,  02 , . . . ,  08 ]  711 – 719 . That is, the interleaver  205  arranges B 1 -S 1 -b 1 , B 2 -S 1 -b 1 , B 3 -S 1 -b 1  and B 4 -S 1 -b 1  in  01 -C 1 , arranges B 1 -S 1 -b 2 , B 2 -S 1 -b 2 , B 3 -S 1 -b 2  and B 4 -S 1 -b 2  in  02 -C 1 , and arranges B 1 -S 1 -b 8 , B 2 -S 1 -b 8 , B 3 -S 1 -b 8  and B 4 -S 1 -b 8  in  08 -C 1 . That is,the interleaver  205  performs interleaving such that the 4×8 Reed-Solomon elements of the first Reed-Solomon symbols S 1  in the first Reed-Solomon block [B 1 ]  311 , the second Reed-Solomon block [B 2 ]  313 , the third Reed-Solomon block [B 3 ]  315  and the fourth Reed-Solomon block [B 4 ]  317  are arranged in first sub-channels of the 8 OFDM symbols. Upon receiving such interleaved signal transmitted by the transmitter, the receiver performs inverse interleaving, i.e., deinterleaving. The deinterleaving means arranging the same sub-channel data blocks of the 8 OFDM symbols in one Reed-Solomon symbol. Therefore, if there exists frequency selective fading or narrow-band jamming signal on the transmission channel, transmission errors occur in a specific sub-channel of the OFDM symbol. The transmission errors occurred in the specific sub-channel are arranged only in a specified one Reed-Solomon symbol by deinterleaving, contributing to an improvement in error correction capability of the Reed-Solomon coding, thereby improving the system performance. 
     The interleaving based on the 16QAM modulation according to the present invention has been described with reference to  FIG. 7 . Next, interleaving based on the 64QAM modulation will be described with reference to  FIG. 8 . 
       FIG. 8  illustrates an OFDM symbol structure and sub-channel arrangement based on 16QAM modulation according to an embodiment of the present invention. 
     Referring to  FIG. 8 , each of OFDM symbols [ 01 ,  02 , . . . ,  08 ]  811 – 819  is comprised of 48 sub-channels [C 1 , C 2 , . . . , C 48 ], and since the 64QAM modulation is used, each sub-channel receives 6-bit data. As described in  FIG. 3 , six Reed-Solomon blocks, e.g., the Reed-Solomon blocks [B 1 , B 2 , B 3 , B 4 , B 5 , B 6 ]  311 – 321  are comprised of 48×8×6 bits. The interleaver  205  arranges the 6 Reed-Solomon blocks [B 1 , B 2 , B 3 , B 4 , B 5 , B 6 ]  311 – 321  in the 8 OFDM symbols [ 01 ,  02 , . . . ,  08 ]  811 – 819  by interleaving. The 8 OFDM symbols  811 – 819  are also comprised of 6×8×48 bits, which are equal to the bit number of 6 Reed-Solomon blocks [B 1 , B 2 , B 3 , B 4 , B 5 , B 6 ]  311 – 321 . Now, a description will be made as to how the interleaver  205  arranges the Reed-Solomon blocks [B 1 , B 2 , B 3 , B 4 , B 5 , B 6 ]  311 – 321  in the 8 OFDM symbols  811 – 819 . 
     The interleaver  205  arranges 8 Reed-Solomon elements [b 1 , b 2 , . . . , b 8 ] of a first Reed-Solomon symbol S 1  of the first Reed-Solomon block [B 1 ]  311 , 8 Reed-Solomon elements [b 1 , b 2 , . . . , b 8 ] of a first Reed-Solomon symbol S 1  of the second Reed-Solomon block [B 2 ]  313 , 8 Reed-Solomon elements [b 1 , b 2 , . . . , b 8 ] of a first Reed-Solomon symbol S 1  of the third Reed-Solomon block [B 3 ]  315 , 8 Reed-Solomon elements [b 1 , b 2 , . . . , b 8 ] of a first Reed-Solomon symbol S 1  of the fourth Reed-Solomon block [B 4 ]  317 , 8 Reed-Solomon elements [b 1 , b 2 , . . . , b 8 ] of a first Reed-Solomon symbol S 1  of the fifth Reed-Solomon block [B 5 ]  319  and 8 Reed-Solomon elements [b 1 , b 2 , . . . , b 8 ] of a first Reed-Solomon symbol S 1  of the sixth Reed-Solomon block [B 6 ]  321  in first sub-channels [ 01 -C 1 ,  02 -C 1 ,  03 -C 1 , . . . ,  08 -C 1 ] of the 8 OFDM symbols [ 01 ,  02 , . . . ,  08 ]  811 – 819 . That is, the interleaver  205  arranges B 1 -S 1 -b 1 , B 2 -S 1 -b 1 , B 3 -S 1 -b 1 , B 4 -S 1 -b 1 , B 5 -S 1 -b 1  and B 6 -S 1 -b 1  in  01 -C 1 , arranges B 1 -S 1 -b 2 , B 2 -S 1 -b 2 , B 3 -S 1 -b 2 , B 4 -S 1 -b 2 , B 5 -S 1 -b 2  and B 6 -S 1 -b 2  in  02 -C 1 , and arranges B 1 -S 1 -b 8 , B 2 -S 1 -b 8 , B 3 -S 1 -b 8 , B 4 -S 1 -b 8 , B 5 -S 1 -b 8  and B 6 -S 1 -b 8  in  08 -C 1 . That is, the interleaver  205  performs interleaving such that the 6×8 Reed-Solomon elements of the first Reed-Solomon symbols S 1  in the first Reed-Solomon block [B 1 ]  311 , the second Reed-Solomon block [B 2 ]  313 , the third Reed-Solomon block [B 3 ]  315 , the fourth Reed-Solomon block [B 4 ]  317 , the fifth Reed-Solomon block [B 5 ]  319  and the sixth Reed-Solomon block [B 4 ]  321  are arranged in first sub-channels of the 8 OFDM symbols. Upon receiving such interleaved signal transmitted by the transmitter, the receiver performs inverse interleaving, i.e., deinterleaving. The deinterleaving means arranging the same sub-channel data blocks of the 8 OFDM symbols in one Reed-Solomon symbol. Therefore, if there exists frequency selective fading or narrow-band jamming signal on the transmission channel, transmission errors occur in a specific sub-channel of the OFDM symbol. The transmission errors occurred in the specific sub-channel are arranged only in a specified one Reed-Solomon symbol by deinterleaving, contributing to an improvement in error correction capability of the Reed-Solomon coding, thereby improving the system performance. 
     As described above, the transmitter interleaves Reed-Solomon coded data symbols according to the present invention through respective sub-channels of the OFDM symbols before transmission, so that when the receiver receives the interleaved OFDM symbols, the errors occurred in the transmission channel exist in only a specified one Reed-Solomon symbol after deinterleaving, thus contributing to an improvement in error correction capability of the Reed-Solomon coding. 
     Next, a sub-channel repetitive transmission scheme according to the present invention will be described with reference to  FIGS. 9 to 11 . 
     The sub-channel repetitive transmission is used in the OFDM system to repeatedly transmit one transmission data block over different OFDM sub-channels. When the sub-channel repetitive transmission is used, the transmission data is resistant to errors occurring in the transmission channel. In addition, since the frequency diversity is acquired by the repetitive transmission, it is possible to provide reliable communication even in a frequency selective fading environment or a poor environment where an intended/non-intended interference signals exist. Further, it is possible to vary the associated sub-channels during the sub-channel repetitive transmission depending on the time. That is, it is possible to acquire additional frequency diversity by varying a frequency of the input data depending on the time. The sub-channel repetitive transmission scheme will be described with reference to  FIGS. 9 to 11 . 
       FIG. 9  illustrates a structure of a sub-channel repeater according to a first embodiment of the present invention. Referring to  FIG. 9 , input data blocks [B( 1 ), B( 2 ), B( 3 ), B( 4 )]  900  are provided to a sub-channel repeater  911 . The sub-channel repeater  911  repeats each of the input data blocks [B( 1 ), B( 2 ), B( 3 ), B( 4 )]  900  over 4 sub-channels. Further, reference numerals  913  of U( 1 ) to U( 16 ) represent sub-channels. Thus, as illustrated in  FIG. 9 , the input data block B 1  is repeated over the sub-channels U( 1 ), U( 5 ), U( 9 ) and U( 13 ), and the input data block B( 2 ) is repeated over the sub-channels U( 2 ), U( 6 ), U( 10 ) and U( 14 ). Further, the input data block B( 3 ) is repeated over the sub-channels U( 3 ), U( 7 ), U( 11 ) and U( 15 ), and the input data block B( 4 ) is repeated over the sub-channels U( 4 ), U( 8 ), U( 12 ) and U( 16 ). As a result, the received input data blocks  900  are subject to sub-channel repetition by the sub-channel repeater  911 , and thus converted to 16 sub-channel data blocks [U( 1 ), U( 2 ), . . . , U( 16 )]  913 . Then, the sub-channel data blocks [U( 1 ), U( 2 ), . . . , U( 16 )]  913  are provided to associated mappers  915  where the provided sub-channel data blocks are subject to mapping for modulation. 
       FIG. 9  shows an example where 4 input data blocks are repeated over 4 sub-channels, and the repeated sub-channel data blocks are mapped in their own unique mappers. However,  FIG. 10  shows an example where the sub-channel data blocks are mapped by grouping. 
       FIG. 10  illustrates a structure of a sub-channel repeater according to a second embodiment of the present invention. Referring to  FIG. 10 , input data blocks  1000 , a sub-channel repeater  1011  and sub-channel data blocks  1013  are identical in function to the input data blocks  900 , the sub-channel repeater  911 , and the sub-channel data blocks  913  of  FIG. 9 . In  FIG. 9 , the sub-channel data blocks  913  are mapped by their associated mappers  915 . In  FIG. 10 , however, the sub-channel data blocks  1013  are mapped by the mappers  1015  by grouping. Here, each of the mappers  1015  maps 4 sub-channels as one modulation symbol. Since the 4 sub-channels having different repeated data blocks are mapped as one modulation symbol, the number of the input data blocks to the sub-channel repeater  1011  and the number of the sub-channels after sub-channel repetition are both equal to 4. Although the 4 sub-channel data blocks are mapped as one modulation symbol in  FIG. 10 , it is also possible to map 2 sub-channel data blocks as one modulation symbol, thereby mapping 8 sub-channels. 
     The sub-channel repetitive transmission scheme has been described with reference to  FIGS. 9 and 10 . Next, an internal structure of the sub-channel repeater for performing the sub-channel repetitive transmission will be described with reference to  FIG. 11 . 
       FIG. 11  illustrates an internal structure of the sub-channel repeater shown in  FIGS. 9 and 10 . Referring to  FIG. 11 , M input data blocks [B( 1 ), B( 2 ), . . . , B(M)]  1100  are provided to a sub-channel repeater  1111 . The sub-channel repeater  1111  then repeats the input data blocks  1100  under the control of a sub-channel repetition controller  1113 . The sub-channel repetition controller  1113  controls the sub-channel repetition using channel information  1115 , and outputs N sub-channel repetition control signals x( 1 ), x( 2 ), x( 3 ), . . . , x(N). The sub-channel repeater  1111  performs sub-channel repetition on the input data blocks  1100  according to the sub-channel repetition control signals output from the sub-channel repetition controller  1113 , and outputs sub-channel data blocks [U( 1 ), U( 2 ), . . . , U(N)]  1119 . In order to specifically describe the sub-channel repetition, a process for converting and outputting the first sub-channel data block U( 1 ) output from the sub-channel repeater  1111  will be described by way of example. The sub-channel repeater  1111  includes N selectors. For example, a first selector  1121  receives the input data blocks  1100  as input data blocks  1123 , selects one of the M input data blocks  1123 , and converts the selected data block to the sub-channel data block U( 1 ). The selector  1121  converts one of the input data blocks  1123  to the sub-channel data block U( 1 ) according to a first sub-channel repetition control signal x( 1 ) output from the sub-channel repetition controller  113 . 
     In  FIGS. 9 to 11 , since the sub-channel repetition transmission scheme according to the second embodiment of the present invention repeatedly transmits one input data block over a plurality of different sub-channels, it is resistant to errors occurring in the transmission channel. In addition, since the frequency diversity is acquired by the repetitive transmission, it is possible to provide reliable communication even in a frequency selective fading environment or a poor environment where an intended/non-intended interference signals exist. Further, it is possible to vary the associated sub-channels during the sub-channel repetitive transmission depending on the time. In this case, it is possible to acquire additional frequency diversity. 
     Next, a scheme for dynamically adaptively assigning sub-channels according to the third embodiment of the present invention will be described with reference to  FIGS. 12A to 13 . 
       FIGS. 12A and 12B  illustrate a sub-channel assignment scheme for frequency transition, especially a scheme for dynamically adaptively performing sub-channel assignment according to an embodiment of the present invention. 
     Referring to  FIG. 12A , sub-channel data blocks [R( 1 ), R( 2 ), . . . , R( 8 )]  1213  applied to a sub-channel assignor  1211  at time t=t1 constitute 8 sub-channels. The received sub-channel data blocks  1213  are dynamically assigned to the associated sub-channels by the sub-channel assignor  1211 , and are output as 8 output sub-channels [A( 1 ), A( 2 ), . . . , A( 8 )]  1215 . For example, at the time t=t 1 , a first input sub-channel data block R( 1 ) is assigned to a third output sub-channel A( 3 ) among the output sub-channels  1215  by the sub-channel assignor  1211 . However, as illustrated in  FIG. 12B , at time t=t 2  after a lapse of time t=t+1, the sub-channel assignor  1211  assigns the input sub-channel data blocks  1213  to the 8 sub-channels [A( 1 ), A( 2 ), . . . , A( 8 )]  1250  in a different manner from the dynamical assignment of  FIG. 12A , i.e., assigns the input sub-channel data blocks  1213  such that frequency transition occurs. That is, the sub-channel assignments are performed differently at time t=t 2  and time t=t 1 . Varying the sub-channel assignment means that transition occurs in terms of a frequency of the sub-channels. Therefore, there occur the effects of the frequency transition of the sub-channels. 
     Next, an internal structure of a sub-channel assignor for controlling the dynamic/adaptive sub-channel assignment described in conjunction with FIGS.  12 A and  12 B will be described with reference to  FIG. 13 . 
       FIG. 13  illustrates an internal structure of a sub-channel assignor according to an embodiment of the present invention. Referring to  FIG. 13 , K input sub-channel data blocks [R( 1 ), R( 2 ), . . . , R(K)]  1311  are provided to a sub-channel assignor  1313 . The sub-channel assignor  1313  then performs dynamic sub-channel assignment on the input sub-channel data blocks  1311 . The dynamic sub-channel assignment by the sub-channel assignor  1313  is performed under the control of a sub-channel assignment controller  1315 . The sub-channel assignment controller  1315  controls the dynamic sub-channel assignment according to channel information  1317 . The sub-channel assignment controller  1315  provides K sub-channel assignment control signals n( 1 ), n( 2 ), n( 3 ), . . . , n(K) to the sub-channel assignor  1313 . The sub-channel assignor  1313  then assigns the input sub-channel data blocks [R( 1 ), R( 2 ), . . . , R(K)]  1311  to the associated output sub-channels [A( 1 ), A( 2 ), . . . , A(K)]  1319  according to the sub-channel assignment control signals n( 1 ), n( 2 ), n( 3 ), . . . , n(K). In order to specifically describe the sub-channel assignment, a process for assigning the input sub-channel data blocks  1311  to a first output sub-channel A( 1 ) among the output sub-channels [A( 1 ), A( 2 ), . . . , A(K)]  1319  will be described by way of example. The input sub-channel data blocks  1311  are converted to input data blocks  1321 . Then, a selector  1323  selects one of the K input sub-channel data blocks  1321  under the control of the sub-channel assignment controller  1315 , and assigns the selected input sub-channel data block to the output sub-channel A( 1 ). Here, the selector  1323  assigns the input sub-channel data block to the corresponding output sub-channel depending on the first sub-channel assignment control signal n( 1 ) generated by the sub-channel assignment controller  1315 . 
     As described with reference to  FIGS. 12A to 13 , the OFDM system performs dynamic sub-channel assignment by varying the sub-channel assignment depending on the time or a specific code pattern, rather than statically assigning the sub-channels, and acquires frequency diversity by adaptively assigning the sub-channels according to the channel conditions, thus contributing to an improvement in system performance. Of course, it is possible to further improve performance in terms of frequency diversity by combining the sub-channel repetitive transmission scheme of the second embodiment with the sub-channel assignment scheme of the third embodiment. 
     Next, a transmission scheme for selecting a minimum PAPR sub-channel without separate supplemental information according to the fourth embodiment of the present invention will be described with reference to  FIGS. 14 to 16 . 
       FIG. 14  illustrates a structure of a minimum PAPR select sub-channel transmitter according to an embodiment of the present invention. It will be assumed that 4 pilot sub-channels [M(p 1 ), M(p 2 ), M(p 3 ), M(p 4 )] are included in K input sub-channels [M( 1 ), M( 2 ), . . . , M(p 1 ), . . . , M(p 2 ), . . . , M(p 3 ), . . . , M(p 4 ), . . . , M(K)]  1411 . Here, the pilot sub-channel transmission points are previously determined in the OFDM system. 
     In a first path, the pilot sub-channel data blocks [M(p 1 ), M(p 2 ), M(p 3 ), M(p 4 )] among the K sub-channel data blocks  1411  are provided to associated multipliers  1413 . The multipliers  1413  multiply the pilot sub-channel data blocks [M(p 1 ), M(p 2 ), M(p 3 ), M(p 4 )] by first pilot scrambling codes [Cp 1 ]  1417  generated by a pilot scrambling code generator  1415 , for phase modulation. For example, the first pilot scrambling codes  1417  have a value of Cp 1 =[1, 1, 1, 1]. The K sub-channel data blocks  1411  including the phase-modulated pilot sub-channel data blocks [M(p 1 ), M(p 2 ), M(p 3 ), M(p 4 )] are scrambled by scramblers  1423  with first scrambling codes [c 1 ( 1 ), c 1 ( 2 ), . . . , c 1 (K)]  1421  generated by a scrambling code generator  1419 . 
     In a second path, the pilot sub-channel data blocks [M(p 1 ), M(p 2 ), M(p 3 ), M(p 4 )] among the K sub-channel data blocks  1411  are provided to associated multipliers  1425 . The multipliers  1425  multiply the pilot sub-channel data blocks [M(p 1 ), M(p 2 ), M(p 3 ), M(p 4 )] by second pilot scrambling codes [Cp 2 ]  1427  generated by the pilot scrambling code generator  1415 , for phase modulation. For example, the second pilot scrambling codes  1427  have a value of Cp 2 =[−1, −1, −1, −1]. The K sub-channel data blocks  1411  including the phase-modulated pilot sub-channel data blocks [M(p 1 ), M(p 2 ), M(p 3 ), M(p 4 )] are scrambled by scramblers  1431  with second scrambling codes [c 2 ( 1 ), c 2 ( 2 ), . . . , c 2 (K)]  1429  generated by the scrambling code generator  1419 . 
     The sub-channel data blocks generated in the first and second paths, i.e., the sub-channel data blocks [S 0 ( 1 ), S 0 ( 2 ), . . . , S 0 (K)] output from the multipliers  1423  and the sub-channel data blocks [S 1 ( 1 ), S 1 ( 2 ), . . . , S 1 (K)] output from the multipliers  1431  are subject to inverse fast Fourier transform by an IFFT  1433  and an IFFT  1435 , respectively. The IFFT-transformed sub-channel data blocks, i.e., the sub-data blocks [s 0 ( 1 ), s 0 ( 2 ), . . . , s 0 (K)]  1437  output from the IFFT  1433  and the sub-data blocks [s 1 ( 1 ), s 1 ( 2 ), . . . , s 1 (K)]  1439  output from the IFFT  1435  are provided to a PAPR calculator  1441  and a PAPR calculator  1443 , respectively. The PAPR calculator  1441  calculates a peak-to-average power ratio PAPR(s 0 ) of the sub-channel data blocks  1437  output from the IFFT  1433 , and provides the PAPR(s 0 ) to a comparator  1445 . Further, the PAPR calculator  1443  calculates a peak-to-average power ratio PAPR(s 1 ) of the sub-channel data blocks  1439  output from the IFFT  1435 , and provides the PAPR(s 1 ) to the comparator  1445 . The comparator  1445  then compares the PAPR(s 0 ) output from the PAPR calculator  1441  with the PAPR(s 1 ) output from the PAPR calculator  1443 , selects a MINIPAPR value  1447  having a lower PAPR, and provides the selected MINIPAPR value  1447  to a selector  1449 . The selector  1449  then selects sub-channel data blocks having a lower PAPR among the sub-channel data blocks  1437  output from the IFFT  1443  and the sub-channel data blocks  1439  output from the IFFT  1435 , based on the MINIPAPR value output from the comparator  1445 , and provides the selected sub-channel data blocks [s( 1 ), s( 2 ), . . . , s(K)] to a P/S converter  1451 . The P/S converter  1451  then converts the parallel input sub-channel data blocks into serial output sub-channel data blocks. Although the embodiment of the present invention has been described with reference to an example where the number of the pilot scrambling codes and the number of the scrambling codes are both 2, the number of the pilot scrambling codes and the number of the scrambling codes are extendable. 
     With reference to  FIGS. 15 and 16 , the present invention will be described regarding an embodiment where the number of the pilot scrambling codes and the number of the scrambling codes are both 4. 
       FIG. 15  illustrates a structure of an extended minimum PAPR select sub-channel transmitter in which the number of IFFTs is extended. Referring to  FIG. 15 , 4 pilot scrambling codes generated to transmit 4 scrambling code information blocks over 4 pilot sub-channels include a first pilot scrambling code Cp 1 =[1, 1, 1, 1], a second pilot scrambling code Cp 2 =[−1, −1, −1, −1], a third pilot scrambling code Cp 3 =[j, j, j, j], and a fourth pilot scrambling code Cp 4 =[−j, −j, −j, −j]. The pilot scrambling codes and the scrambling codes are previously recognized by both the transmitter and the receiver. 
     A pilot sub-channel data generator  1511  generates transmission pilot sub-channel data  1513 . It will be assumed herein that the transmission pilot sub-channel data  1513  has a value of [1, 1, 1, −1]. A pilot scrambling code generator  1515  generates the first pilot scrambling code Cp 1 =[1, 1, 1, 1], the second pilot scrambling code Cp 2 =[−1, −1, −1, −1], the third pilot scrambling code Cp 3 =[j, j, j, j], and the fourth pilot scrambling code Cp 4 =[−j, −j, −j, −j]. Multipliers  1517 ,  1519 ,  1521  and  1523  multiply the pilot sub-channel data  1513  generated by the pilot sub-channel data generator  1511  by the first pilot scrambling code Cp 1 =[1, 1, 1, 1], the second pilot scrambling code Cp 2 =[−1, −1, −1, −1], the third pilot scrambling code Cp 3 =[j, j, j, j], and the fourth pilot scrambling code Cp 4 =[−j, −j, −j, −j], respectively. The multiplier  1517  outputs a signal of [1, 1, 1, −1], the multiplier  1519  outputs a signal of [−1, 31 1, −1, 1], the multiplier  1521  outputs a signal of [j, j, j, −j], and the multiplier  1523  outputs a signal of [−j, −j, −j, j]. The scrambled pilot sub-channel data blocks output from the multipliers  1517 ,  1519 ,  1521  and  1523  are added to data  1535  on the sub-channel transmitting actual data by pilot adders  1525 ,  1527 ,  1529  and  1531 , respectively. The signals output from the pilot adders  1525 ,  1527 ,  1529  and  1531  are scrambled by scramblers  1537 ,  1539 ,  1541  and  1543  with scrambling codes  1547  generated by a scrambling code generator  1545 . The scrambling codes  1547  generated by the scrambling code generator  1545  include a first scrambling code Cd 1 , a second scrambling code Cd 2 , a third scrambling code Cd 3  and a fourth scrambling code Cd 4 . The signals output from the scramblers  1537 ,  1539 ,  1541  and  1543  are provided to IFFTs  1549 ,  1551 ,  1553  and  1555 , respectively. The IFFTs  1549 ,  1551 ,  1553  and  1555  IFFT-transform the signals output from the scramblers  1537 ,  1539 ,  1541  and  1543 , respectively, and provide their outputs to PAPR calculators  1557 ,  1559 ,  1561  and  1563 . The PAPR calculators  1557 ,  1559 ,  1561  and  1563  then calculate PAPRs of the signals provided from the IFFTs  1549 ,  1551 ,  1553  and  1555 , respectively, and provide their outputs to a PAPR comparator &amp; minimum PAPR selector  1565 . The PAPR comparator &amp; minimum PAPR selector  1565  compares the PAPRs of the sub-channel data blocks calculated by the PAPR calculators  1557 ,  1559 ,  1561  and  1563 , selects a sub-channel data block having a minimum PAPR, and transmits the selected sub-channel data over the sub-channel. 
     If it is assumed in  FIG. 15  that the sub-channel data block scrambled by the third scrambling code Cd 3  has the minimum PAPR, the pilot sub-channel data [1, 1, 1, −1]  1513  is scrambled with the third scrambling code Cp 3 =[j, j, j, j,], generating the scrambled pilot sub-channel data block [j, j, j, −j]. For the sake of convenience, if it is assumed that the scrambling code where the pilot channel exists is [1, 1, 1, 1] (of course, [j, 1, 1, j] is also available), the transmission pilot sub-channel data is transmitted in the form of [j, j, j, −j]. 
     Next, a structure of a receiver corresponding to the minimum PAPR select sub-channel transmitter described in conjunction with  FIG. 15  will be described with reference to  FIG. 16 . 
       FIG. 16  illustrates a structure of a receiver corresponding to the minimum PAPR select sub-channel transmitter of  FIG. 15 . Referring to  FIG. 16 , a signal received over a radio channel is provided to a frequency synchronization acquirer  1611 . The frequency synchronization acquirer  1611  acquires synchronization between the transmitter and the receiver by performing rough frequency synchronization and fine frequency synchronization, and provides the frequency-synchronized channel data to a fast Fourier transformer (FFT)  1613 . The FFT  1613  then FFT-transforms the channel data output from the frequency synchronization acquirer  1611 , and provides its output to a channel estimator and equalizer  1615 . The channel estimator and equalizer  1615  performs channel estimation and equalization on the signal provided from the FFT  1613 . The data output from the channel estimator and equalizer  1615  is provided to a pilot extractor  1617 . The pilot extractor  1617  extracts pilot sub-channel data from the output data of the channel estimator and equalizer  1615 . Here, the position where the pilot sub-channel data of the received channel signal exists is previously agreed by the transmitter and the receiver. 
     A scrambling code generator  1619  generates the same scrambling codes as used by the transmitter, i.e., generates a first scrambling code Cd 1 , a second scrambling code Cd 2 , a third scrambling code Cd 3  and a fourth scrambling code Cd 4 . The generated scrambling codes are provided to associated pilot extractors  1621 . The pilot extractors  1621  extract pilot channel data blocks Cdp 1 , Cdp 2 , Cdp 3  and Cdp 4 , respectively, and provide the extracted pilot channel data blocks to associated complex conjugate operators  1623 . The complex conjugate operators  1623  complex-conjugate the extracted pilot channel data blocks Cdp 1 , Cdp 2 , Cdp 3  and Cdp 4 , respectively. The signals output from the complex conjugate operators  1623  are multiplied by multipliers  1625  by the signal output from the pilot extractor  1617 , generating pilot sub-channel data blocks [j, j, j, −j], [j, j, j, −j], [j, j, j, −j] and [j, j, j, −j] in which the effects of the scrambling codes are removed. 
     A pilot scrambling code generator  1627  also generates the same pilot scrambling codes as used by the transmitter, i.e., generates a first pilot scrambling code Cp 1 =[1, 1, 1, 1,], a second pilot scrambling code Cp 2 =[−1, −1, −1, −1], a third pilot scrambling code Cp 3 =[j, j, j, j], and a fourth pilot scrambling code Cp 4 =[−j, −j, −j, −j]. The 4 pilot scrambling codes generated by the pilot scrambling code generator  1627  are provided to associated complex conjugate operators  1629 . The complex conjugate operators  1629  complex-conjugate the pilot scrambling codes, and provides the complex-conjugated pilot scrambling codes Cp 1 *=[1, 1, 1, 1], Cp 2 *=[−1, −1, −1, −1], Cp 3 *=[−j, −j, −j, −j] and Cp 4 *=[j, j, j, j] to associated multipliers  1631 . The multipliers  1631  multiply the signals output form the multipliers  1625  by the signals output from the complex conjugate operators  1629 , and generate signals [j, j, j, −j], [−j, −j, −j, j], [1, 1, 1, −1] and [−1, −1, −1, 1] in which even the effects of the pilot scrambling codes are removed. 
     A pilot sub-channel data generator  1635  generates the same pilot sub-channel data as generated by the transmitter, i.e., generates pilot sub-channel data [1, 1, 1, −1], and provides the generated pilot sub-channel data to a complex conjugate operator  1637 . The complex conjugate operator  1637  complex-conjugates the pilot sub-channel data [1, 1, 1, −1]. The complex-conjugated pilot sub-channel data [1, 1, 1, −1] is provided to multipliers  1639 . The multipliers  1639  multiply the complex-conjugated pilot sub-channel data by the data blocks output from the associated multipliers  1631 , and generate signals [j, j, j, j], [−j, −j, −j, −j], [1, 1, 1, 1] and [−1, −1, −1, −1] in which even the effects of the pilot sub-channel data are completely removed. 
     As described in conjunction with  FIG. 16 , when the 4 pilot scrambling codes are used, 4 elements of each of the finally processed pilot sub-channel data blocks [j, j, j, j], [−j, −j, −j, −j], [1, 1, 1, 1] and [−1, −1, −1, −1] have the same values, and the 4 signals have a phase difference of 90 degree from one another. When the transmitter performs scrambling using 4 scrambling codes, selects only a specific sub-channel data block having the minimum PAPR and transmits the selected sub-channel data block, a branch of the specific scrambling code used has a value of [1, 1, 1, 1]. Therefore, the receiver can identify the scrambling code used by the transmitter by determining a branch closest to [1, 1, 1, 1] using the 4 signals. In  FIG. 16 , since the branch closest to [1, 1, 1, 1] is the third sub-channel data block, the receiver can recognize that the transmitter performed scrambling using the third scrambling code Cd 3 . A decider and scrambling code information detector  1641  determines the scrambling code used by the transmitter, as described above. When the decider and scrambling code information detector  1641  determines the scrambling code used by the transmitter, a scrambling code generator  1643  selects a scrambling code among the scrambling codes generated by the scrambling code generator  1619  based on the scrambling code information detected by the decider and scrambling code information detector  1641 , and provides the selected scrambling code to a multiplier  1645 . The multiplier  1645  multiplies the selected scrambling code by the signal output from the channel estimator and equalizer  1615 , and provides its output to a demodulator  1647 . The demodulator  1647  receives the data output from the multiplier  1645  and demodulates the received data into original data transmitted by the transmitter. 
     As described in conjunction with  FIGS. 14 to 16 , in order to reduce the PAPR, the OFDM system scrambles sub-channel data blocks using a plurality of scrambling codes, IFFT-transforms the scrambled sub-channel data blocks, selects a sub-channel data block with the minimum PAPR, and transmits the selected sub-channel data block. Hence, the receiver can recognize the scrambling code used by the transmitter by demodulating a plurality of pilot sub-channels, even though the transmitter has not transmitted separate supplemental information on the scrambling code. Therefore, the fourth embodiment of the present invention need not transmit the separate supplemental information, it is possible to maintain the transmission efficiency. Further, the receiver can extract scrambling code information without performing demodulation on the supplemental information, thus contributing to simplification of the hardware structure of the receiver. 
     Next, a transmission antenna diversity scheme according to the fifth embodiment of the present invention will be described in detail with reference to  FIG. 17 . 
       FIG. 17  illustrates a transmission diversity scheme according to an embodiment of the present invention. Referring to  FIG. 17 , an input signal x(t)  1701  is transmitted over two paths. A signal  1702  transmitted over the first path has the same phase as the input signal x(t)  1701  (i.e., has an offset of a zero degree phase  1704 ), and is transmitted as a transmission signal x 1 (t)  1710  through a first transmission antenna  1708 . A signal  1703  transmitted over the second path is further transmitted over two sub-paths: one signal transmitted over a first sub-path has an in-phase offset of a zero degree phase  1705 , and another signal transmitted over a second sub-path has a phase-inversed offset of a  180  degree phase  1706 . The signal having the in-phase offset  1705  and the signal having the phase-inversed offset  1706  are alternately selected by a switch  1707  in training symbol period. The signal selected by the switch  1707  is transmitted as transmission signal x 2 (t)  1711  through a second transmission antenna  1709 . The output signal x 1 (t)  1710  of the first transmission antenna  1708  is received at reception antenna  1714  through a first transmission path h 1 (t)  1712 , and the output signal x 2 (t)  1711  of the second transmission antenna  1709  is received at the reception antenna  1714  through a second transmission path h 2 (t)  1713 . An output signal r(t) of the reception antenna  1714  is provided to a reception signal processor  1715 . The reception signal processor  1715  performs channel estimation and channel compensation on the two transmission paths, and then performs data demodulation. 
     A detailed description of the transmission diversity scheme will be made herein below. The signals x 1 (t)  1710  and x 2 (t)  1711  transmitted through the first and second transmission antennas  1708  and  1709  at time t=t 1  and time t=t 2 , are defined as
 
 x 1( t )= x ( t ) at time  t=t 1
 
 x 1( t )= x ( t ) at time  t=t 2
 
 x 2( t )= x ( t ) at time  t=t 1
 
 x 2( t )=− x ( t ) at time  t=t 2
 
     Further, the signals received at the receiver are defined as
 
 r ( t )= h 1( t )* x 1( t )+ h 2( t )* x 2( t ) at time  t=t 1  (1)
 
 r ( t )= h 1( t )* x 1( t )+ h 2( t )*(− x 2( t )) at time  t=t 2  (2)
 
     In Equations (1) and (2), “*” denotes convolution. If it is assumed that the transmitter transmits training symbols in a training symbol period for channel estimation on a transmission frame, the signals x(t 1 ) and x(t 2 ) at time t=t 1  and time t=t 2  are equal to each other. 
     That is, in the training symbol period, Equations (1) and (2) are expressed as Equations (3) and (4).
 
 r   t1,tr ( t )= h 1( t )* x   tr ( t )+ h 2( t )* x   tr ( t )  (3)
 
 r   t2,tr ( t )= h 1( t )* x   tr ( t )− h 2( t )* x   tr ( t )  (4)
 
     In Equations (3) and (4), the training symbols received at time t=t 1  and time t=t 2  are the signals transmitted in the training symbol period. 
     Therefore, Equations (5) and (6) represent transfer functions calculated using Equations (3) and (4).
 
 R   t1,tr =( H 1+ H 2) X   tr   (5)
 
 R   t2,tr =( H 1− H 2) X   tr   (6)
 
     Hence, transfer functions for the transmission channels over the two paths can be calculated as follows using Equations (5) and (6). 
     
       
         
           
             
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     Therefore, it is possible to improve the system performance by applying the determined characteristics of the 2 transmission channels to the data symbols received after the training symbol period. As a result, it is possible to estimate the channels over the transmission paths transmitted by the transmitter through 2 transmission antennas by utilizing the transmission antenna diversity scheme according to the fifth embodiment of the present invention. Accordingly, it is possible to improve system performance by processing and demodulating data using the estimation results on the 2 transmission channels. 
     The present invention has the following advantages. 
     First, the first embodiment interleaves/deinterleaves data symbols such that a group of error (or damaged) data blocks on an OFDM transmission channel is arranged in a specified one of Reed-Solomon coded symbols. That is, this embodiment improves error correction capability for the frequency selective fading by performing interleaving and deinterleaving such that respective data blocks in one Reed-Solomon symbol should be mapped to the same sub-channels in a plurality of OFDM symbols. 
     Second, the second embodiment acquires frequency diversity by performing repetitive transmission on a plurality of different OFDM sub-channels in the OFDM system. Hence, the OFDM system provides reliable data communication even in a frequency selective fading environment or a poor environment where an intended/non-intended interference signals exist. Further, it is possible to perform channel mapping such that during repetitive transmission, the associated sub-channels vary depending on the time, thus acquiring additional frequency diversity. 
     Third, the third embodiment dynamically performs OFDM sub-channel assignment according to a predetermined code pattern or a pattern previously set in the OFDM system depending on the time, rather than statically performing sub-channel mapping, or adaptively performs the sub-channel assignment according to the channel condition. Since the sub-channel frequency is not static but dynamic, it is possible to acquire frequency diversity. 
     Fourth, in an OFDM system according to the fourth embodiment, a receiver detects a selected sub-channel with the minimized PAPR (Peak-to-Average Power Ratio) using a plurality of scrambling codes, even though a transmitter does not transmit separate supplemental information. The minimization of the PAPR reduces a load of a power amplifier (PA) in the transmitter, making it possible to readily implement the power amplifier. In addition, even though the transmitter does not transmit the supplemental information for the scrambling code, the receiver can detect the sub-channel selected by the transmitter through the pilot sub-channel, thus contributing to simplification of the hardware structure of the transceiver. 
     Fifth, the fifth embodiment implements transmission antenna diversity for alternating phases in a training symbol period so that the receiver can estimate the characteristics of different transmission channels when diversity is applied to the transmission antennas in the OFDM system. Accordingly, the receiver can perform channel estimation on the respective transmission paths used by the transmitter in transmitting the signals through two antennas, and performs data processing and demodulation using the estimation results on the respective transmission channels, thus improving system performance. 
     While the invention has been shown and described with reference to a certain preferred embodiment thereof, it will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the spirit and scope of the invention as defined by the appended claims.