Abstract:
A method for operating an alternating-current (AC) controller system includes providing a first bi-directional switch coupled to a load and an AC power source. The first bi-directional switch is a solid-state device. The first switch is turned on in a first half-cycle of an AC cycle. The first switch is turned off in the first half-cycle of the AC cycle.

Description:
CROSS-REFERENCES TO RELATED APPLICATIONS 
     This application claims priority to U.S. Provisional Patent Application No. 60/343,743, filed on Oct. 31, 2001, which is incorporated by reference herein for all purposes. 
    
    
     BACKGROUND OF THE INVENTION 
     FIG. 1A shows a conventional AC controller  200  having a silicon control rectifier (“SCR”) as the solid-state device switch. The controller  200  includes a mains or power source  202  that supplies power, a switch  202  that regulates the power, and a control circuit  204  that controls the turn on and off characteristics of the switch. A load  206  receives the power. 
     The switch  202  includes a first silicon control rectifier (“SCR”)  208  and a second SCR  210  that are arranged in an “anti-parallel” formation to conduct currents in both directions. Like a diode, an SCR generally conducts currents in a single direction so two SCRs are provided in a reverse orientation to serve as an AC switch. 
     There are three basic control modes in SCR devices: (1) on/off, (2) zero-firing, and (3) phase-firing. The first mode or on/off mode is the simplest method and replicates the action of an electromechanical switch. The power is either turned on or turned off according to the commands of the control circuits  204 . Generally, the device is “on” if a command signal is applied to the SCR, and the device is “off” if the command signal is removed. 
     The second control mode or zero-firing mode switches the SCRs on and off, but provides a proportional control capability. With this control mode, the number of “on” or “off” AC cycles is varied to maintain a steady voltage level to the load while turning the power on and off. While effective, the zero-firing is not suited for some application. The voltage applied to the load is either zero or full because zero-firing turns the SCR either completely on or off. This is not suitable for certain exotic load elements, such as, molybdenum disilicide. Molybdenum disilicide&#39;s resistance is nearly zero when cold, but it increases with temperature. A large current surge results each time the SCR is turned on from a cold state. These current surges can damage SCRs. 
     The third control mode or phase-firing provides infinite variable control of voltage being applied to the load, much like a light dimmer. Similar to the zero-firing, the phase firing provides timed gate pulses or command signals to the SCRs. The phase-firing mode, however, turns on each of the two SCRs in an AC switch only for a portion of the respective half-cycles. 
     Referring to FIG. 1B, the SCRs are being fired on in the AC cycle at a given angle α, as described in the current and voltage waveforms. Once fired on, as long as there is a forward-on current flow iA 1  (FIG.  1 A), the SCR stays on. The SCR turns off as soon as the current iA 1  decreases to substantially zero current level or below the threshold current level of the SCR. As illustrated, the voltage and current waveform is a function of the firing angle α. 
     FIGS. 2A-2C illustrate the current and voltage waveforms of the AC controller  200  as a function of the firing angle α for a resistive load (FIG.  2 A), for a resistive-inductive load (FIG.  2 B), and inductive load (FIG.  2 C). 
     Even at resistive loads, a firing angle α&gt;0, indicating that power consumption of the load is controlled by the AC controller, leads to the generation of first harmonic reactive power and of further harmonic content caused by the distortion of the current waveform. This creates high electromagnetic noise or “EMC” for the AC controller. A countermeasure step, therefore, is required to compensate the reactive power component to reduce the EMC. The current flow at firing angles α&gt;0, is intermittent on the mains side and on the load side. This also causes an increase of EMC in the AC controller  200 . 
     Although the load and firing angle α determine the current waveform, there is no mechanism in the AC control circuit  200  to control them. The EMC generally is reduced in such circuits by adding passive filters. These added filters add to the cost, size and weight of the AC control circuits. 
     SUMMARY OF THE INVENTION 
     Embodiments of the present invention provides the capability of an AC switch to be turned on and off in any time during the AC cycle, allowing the user to apply any desirable pulse pattern to said AC switches, even turning them on and off multiple times within the AC cycle. Accordingly, inverters and converters can be developed with enhanced control features that reduce undesirable noise problems, improve dynamic response of the system to interference or to changing power demands, and improve power regulation and the efficiency of the system. 
     In one embodiment, a method for operating an alternating-current (AC) controller system includes providing a first bi-directional switch coupled to a load and an AC power source. The first bi-directional switch is a solid-state device. The first switch is turned on in a first half-cycle of an AC cycle. The first switch is turned off in the first half-cycle of the AC cycle. 
     In another embodiment, an alternating-current (AC) controller system includes a first switch including a reverse blocking insulated gate bipolar transistor (“RIGBT”) coupled to a power supply to regulate a current supplied by the power supply. The first switch is configured to be turned off while the current is flowing through the first RIGBT. 
     In another embodiment, a method for operating an alternating-current (AC) controller system including providing an AC switch coupled to a power supply and a load. The AC switch is turned on to supply a current to the load. The AC switch is turned off while the current is flowing through the switch and being supplied to the load. 
     In another embodiment, an AC controller includes an AC source having a first pole and a second pole, a load having a first node and a second node, and a first bidirectional switch, a solid state device, that is coupled to the first pole of the AC source and the first node of the load. The bidirectional switch has at least one reverse blocking insulated gate bipolar transistor (IGBT). 
     In yet another embodiment, a multi-phase switch system includes a first AC controller including an AC source having a first pole and a second pole, a load having a first node and a second node, and a first bidirectional switch being a solid state device coupled to the first pole of the AC source and the first node of the load. The bidirectional switch has at least one reverse blocking insulated gate bipolar transistor (IGBT). The system also includes a second AC controller including a second AC source having a first pole and a second pole, a second load having a first node and a second node, and a second bidirectional switch being a solid state device coupled to the first pole of the AC source and the first node of the load. The second bidirectional switch has at least one reverse blocking IGBT. 
     In yet another embodiment, a method for operating an AC controller includes providing an AC source having a first pole and a second pole; providing a first load having a first node and a second node; providing a first bidirectional switch being a solid state device coupled to the first pole of the AC source and the first node of the first load, wherein the bidirectional switch has at least one reverse blocking IGBT; and controlling the first switch to adjust a power factor for optimal performance of the AC controller with respect to the AC source. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1A shows a conventional AC controller having a SCR switch. 
     FIG. 1B shows a voltage level and current flow in the AC controller of FIG.  1 A. 
     FIG. 2A shows a voltage level and current flow for a conventional AC controller coupled to a resistive load. 
     FIG. 2B shows a voltage level and current flow for a conventional AC controller coupled to a resistive-inductive load. 
     FIG. 2C shows a voltage level and current flow for a conventional AC controller coupled to an inductive load. 
     FIG. 3A shows an AC controller having a bi-directional switch according to one embodiment of the present invention. 
     FIG. 3B shows a voltage level and current flows associated with the AC controller of FIG.  3 A. 
     FIG. 4A shows an AC controller having a bi-directional switch according to one embodiment of the present invention. 
     FIG. 4B shows a voltage level and current flows associated with the AC controller of FIG.  4 A. 
     FIGS. 5-8 are schematic views illustrating the fabrication of a reverse blocking IGBT used in the AC switch according to one embodiment of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     In one embodiment, an AC switch or controller includes a reverse blocking insulated gate bipolar transistor (“reverse blocking IGBT,” or “RIGBT”). 
     FIG. 3A illustrates an AC controller  300  configured for resistive loads and/or resistive-capacitive loads without the need of an additional load voltage source (the load should not behave like an inductance). The AC controller  300  includes a power source or mains  302 , a switch  304 , a capacitor  305  (Cn), and an inductor  306 . A RC load  308  receives the currents from the mains  302  according to the controls of the switch  304 . The switch  304  is a solid device, e.g., RIGBT, that is capable of handling currents in two directions unlike the SCRs in the controller  100  of the conventional technology. In one embodiment, the switch  304  includes a first RIGBT T 1  and a second RIGBT T 2  that are arranged in an anti-parallel arrangement. The RIGBTs T 1  and T 2  may have a single or a plurality of dice in a parallel arrangement. The capacitor  305  and the inductor  306  comprise an input filter. The capacitor  305  suppresses voltage spikes across the switch  304  during turn offs. The inductor  306  facilitates reduction in reactive power consumption and current ripple. 
     FIG. 3B shows a mode of operation of the AC controller  300  accordingly to one embodiment of the present invention. The AC controller  300  having the RIGBT switch  304  provides various control schemes for optimal operation because the RIGBT switch can be turned on or off at any point during the AC power cycle. A graph  320  illustrates the voltage level and current flow with respect to a given time, i.e., one cycle. A graph  322  illustrates the turn-on and turn-off states of the switch  304  according to the current level. A graph  324  illustrates the turn-on and turn-off pulse patterns-corresponding to the graph  322 . The first RIGBT T 1  is turn on in the first half-cycle, and the second RIGBT T 2  is turn on in the second half-cycle. As shown, the switch is turned on and off many times in each half AC cycle. 
     First, the AC controller  300  provides capability of adjusting the power factor of the mains  302 . The power factor is the ratio of active power to apparent power. Generally, the power factor is deemed to be cos (θ) for sinusoidal voltage and current, such as alternating current (AC), when the phase difference between the voltage and current is θ. Accordingly, the power factor should be close to one for optimal performance of the circuit. In the AC controller  300 , the power factor can be adjusted to be substantially one, which corresponds to resistive load behavior since the switch can be turn on or off any time during the current cycle (see, FIG. 3 b ). Furthermore, it may be adjusted differently, which offers the possibility to actively compensate reactive power generated elsewhere in the grid. 
     Second, the current flowing through the mains  300  may be controlled to be continuous, driven by the inductance of the mains or the inductor  306  connected in series with the mains  302  to avoid EMC emission problems. In one embodiment, another inductor may be serially connected to the inductor  306 . 
     Third, the AC controller  300  can be controlled to vary circuit conditions faster since the switch  304 , which is self commutated, can be turned off at any time, whenever it is necessary. Accordingly, there is no need to wait for a zero current condition to effectuate the turn off the AC controller  300 , as is the case with the AC controller  200  with an SCR. 
     FIG. 4A shows an AC controller  400  configured for loads with an inductive behavior. The AC controller  400  includes a first voltage source or mains  402 , a second voltage source or mains  404 , a first RIGBT switch  408 , a second RIGBT switch  410 , a capacitor  411  (Cn) and an inductor  412 . A load  406  receives the currents supplied by the mains  402  according to the controls of the first switch. The load  406  includes a resistor and an inductor, as shown in FIG.  4 A. The first mains  402  provides greater power than the second mains  404  according to one embodiment of the present invention. In another embodiment, the first mains  402  and the second mains  402  provide substantially equal power. In yet another embodiment, the second mains  404  provides greater power than the first mains  402 . The first switch  408  includes a first RIGBT T 1  and a second RIGBT T 2 , and the second switch  410  includes a third RIGBT T 3  and a fourth RIGBT T 4  according to one embodiment of the present invention. The second switch  410  provides a free wheeling path to prevent the first switch  408  from being damaged during the turn off of the inductive current. The capacitor  411  and the inductor  412  comprise an input filter. The capacitor  411  suppresses voltage spikes across the switch  408  during turn offs. The inductor  412  facilitates reduction in reactive power consumption and current ripple. 
     FIG. 4B shows a control method of the AC controller  400  according to one embodiment of the present invention. A graph  420  illustrates a voltage level and current flow with respect to a given time, i.e., one cycle. A graph  422  illustrates first, second, and third current flows in the AC controller  400 . The first current i 12  flows from the first switch  408  to the load  406 . The second current iL flows through the load  406 . The third current i 34  flows from the second switch  410  to a node provided between the first switch  408  and the load  406 . A graph  424  shows the turn-on and turn-off states of the first and second switches  408  and  410 . As shown, the switch is turned on and off many times in each half AC cycle. 
     In one embodiment, the first RIGBT T 1  of the first switch  408  and the third RIGBT T 3  of the second switch  410  are turn together. The first RIGBT T 1  is turned on in a pulse pattern during much of the first half of the current cycle, while the third RIGBT T 3  remains turned on during this period. On the other hand, the second RIGBT T 2  is turned on in a pulse pattern during much of the second half of the current cycle, while the fourth RIGBT T 4  remains turned on during this period. 
     The AC controller  400  provides forward conduction capability to the unidirectional free wheeling RGIBTs T 3  or T 4  (the switch  410 ) as long as it might carry load current according to the latter&#39;s polarity. The reverse blocking IGBT functions as a unidirectional free wheeling switch because it will maintain a reverse blocking capability while the gate is turned on. 
     Referring to FIGS. 4A and 4B, the operation of the AC controller  400  is provided as follows. The operation with the first switch  408  continuously closed corresponds to an SCR AC controller with “firing angle” α=0. The AC controller  400  including the reverse blocking IGBT provides additional controllability. An unused part of mains voltage-time-area may be used for ‘earlier’ magnetization or de-magnetization of the load inductance, respectively. This capability contributes to a reduction of first harmonic reactive power. Thus the control towards lower load currents does not lead to an increase of reactive power consumption as in the conventional AC controller. Rather, it instead may be used to reduce the reactive power. 
     As already explained above, the mains current may be controlled to be continuous, driven by mains inductance or the series connection of mains inductance and the inductor  412  to prevent EMC emission problems. The load current flow may also be continuous with the same effect. 
     The AC controller  400  can be turned off any time because the first switch  408  can be turned off at any time during the cycle, as desired. That is, there is no need to wait for zero current turn off as in the conventional SCR based circuits. 
     FIGS. 5-8 illustrate a method of fabrication for a reverse blocking IGBT used in the AC switch according to one embodiment of the present invention. The present fabrication method begins with a semiconductor substrate such as an N+ type substrate  101  and the like of FIG.  5 . It should be noted that the present fabrication method relies upon an N+ type substrate, but may also use other types of substrates. The N+ type substrate includes an N− type layer  103  defined thereon by way of standard chemical vapor deposition (CVD) techniques and the like. The N− type layer includes an N type impurity such as phosphorous or the like at a concentration ranging from about 10 13  atoms/cm 3  to 10 14  atoms/cm 3 , and is preferably at about 4×10 13  atoms/cm 3  for preferred bipolar transistor operation. Relative to the N− type layer, the N+ type semiconductor substrate includes an N type impurity such as phosphorous or the like at a concentration ranging from about 10 15  atoms/cm 3  to about 10 19  atoms/cm 3 , and is preferably at about 10 17  atoms/cm 3 . Of course, other concentrations may also be provided depending upon the particular application. 
     Active IGBT devices define onto the N− type layer by way of, for example, a double diffused MOS (DMOS) technique and others. The DMOS technique defines a gate electrode layer  109  overlying a thin layer of high quality oxide  111 . The gate electrode layer is typical made of polysilicon and the like, which is preferably doped with an N type dopant material for conductivity. Steps of masking and etching define the gate electrodes (G) overlying the thin high quality oxide formed over the N− type layer. Also shown are field plate layers formed overlying a portion of the N− type layer. 
     An implant step(s) forms P type well regions  105  in the N− type layer as illustrated by FIG.  7 . Each P type well region is preferably a P/P+ type well or the like, and is defined between each of the gate electrodes. The P type well region includes a boron impurity concentration ranging from about 10 14  atoms/cm 3  to about 10 18  atoms/cm 3 , and is preferably at about 10 16  atoms/cm 3 . The implant step also forms P type guard ring region(s)  115 . The P type guard ring regions are defined at an outer periphery of the active cell region for the purpose of preventing the conductive region of forming outside the main junction region. Thus, the P type guard ring regions preserve the high voltage characteristics of the present IGBT device. A P type region  116  defining a drain region (D) is formed overlying the backside of the N+ type semiconductor substrate in an implant step. The P type region includes a boron impurity concentration ranging from about 10 15  atoms/cm 3  to about 10 18  atoms/cm 3 , and is preferably at about 10 18  atoms/cm 3 . A subsequent diffusion step creates the P type drain region, which can range in depth from about 50 microns to about 300 microns, and is preferably at about 100 microns for a 600 volt to 3,000 volt IGBT device. The P type impurity for the P type well region, the P type guard ring region, and the P type drain region is preferably boron or the like. 
     A P type region  701  is also defined at the scribe line of the integrated circuit chip. A P type region  107  is also defined from the backside of the wafer. Both of the P type regions are defined by way of sputtering, implantation or the like using an impurity with a higher mobility than, for example, the P type well region, the P type guard ring region, and the P type drain region. By way of a subsequent diffusion step(s), the P type regions  701 ,  107  diffuse faster than the P type impurities of, for example, the well region, the guard ring region, and the drain region. The faster diffusion rate allows the P type regions to connect to each other  117 . This forms a continuous P type “frame” (or diffusion region) around the periphery of the integrated circuit, thereby eliminating the N+ /P+ junction region of the conventional IGBT device. The P type impurity with the higher mobility is preferably aluminum or the like. A step of selective sputtering coats selected regions of the integrated circuit with the aluminum for subsequent thermal diffusion or the like. 
     A source implant step forms an N type source region(s)  107  (S) within the periphery of the P type well region(s)  105 . The source implant is preferably an arsenic implant where the arsenic is at a concentration ranging from about 10 17  atoms/cm 3  to about 10 20  atoms/cm 3 , and is preferably at about 3×10 19  atoms/cm 3 . A metallization layer typically aluminum or the like defines a source metallization layer. As shown, the source (S), the gate (G), and the drain (D) define the IGBT according to the present invention. 
     Optionally, an N+ type dopant  704  such as phosphorous or the like forms selected N+type regions in the drain region. The N+ type regions modify the present IGBT device performance for special switching and forward voltage drop characteristics. The N+ type regions include a phosphorous impurity at a concentration ranging from about 10 16  atoms/cm 3  to about 10 19  atoms/cm 3 , and are preferably at about 7×10 18  atoms/cm 3 . 
     The above detailed descriptions are provided to illustrate specific embodiments of the present invention and are not intended to be limiting. For example, the AC controller described herein can be readily applied to three phase systems also with the similar principles. Those skilled in the art can easily modify the description provided above in connection with a single-phase system to more than one phase systems. Numerous modifications and variations within the scope of the present invention are possible. Accordingly, the present invention is defined by the appended claims.