Abstract:
Systems and methods can provide an improved broadband linearizer that includes a distortion generator with a bypass path for generating both composite triple beat (CTB) and composite second order (CSO) distortions suitable for linearizing a laser. The linearized laser can be suitable for injection into a communications network such as, for example, a hybrid fiber coaxial (HFC) network, among others.

Description:
RELATED APPLICATIONS 
     This application claims priority as a non-provisional of U.S. Provisional Patent Application Ser. No. 61/437,440, entitled “Broadband Linearizer with Combined Second and Third Order Generation with Adjustable Tilt,” filed Jan. 28, 2011, which is incorporated herein by reference in its entirety. 
    
    
     TECHNICAL FIELD 
     This disclosure relates to a broadband linearizer where distortion amplitude and phase matching is performed over a frequency range. 
     BACKGROUND 
     Linearizers can be used in broadband communication equipment where linearization is desired over a wide frequency range. The linearization can perform distortion amplitude and phase matching over a wide frequency range. Generally the distortion amplitude and phase follow a smooth function of frequency. 
     WIPO Application No. PCT/US2006/023641 (WIPO Application), which is incorporated by reference in its entirety herein, teaches a full quadrant linearizer that can cover wide amplitude and phase range for composite second order distortion (CSO). The WIPO Application discloses in FIG. 2 a linearizer with an in-phase distortion generation function and a quadrature distortion generation. The distortion generator can generate frequency independent second order distortion with positive or negative phase (in-phase) and frequency dependent second order components that have a quadrature phase. A frequency dependent in-phase component can be desired and would require modification to the distortion generator disclosed in FIG. 2 of the WIPO Application. Generation of second and third order distortion can be desired, and can be done with a separate second and third order distortion generation path. 
     U.S. Pat. No. 5,132,639, (&#39;639 patent) which is incorporated by reference in its entirety herein, teaches generation of second and third order distortion (CTB or composite triple beat) in separate bypass paths that are recombined into a main signal path. The &#39;639 patent discloses a distortion generator in FIG. 6 with one path for second order distortion generation that includes a distortion generator, a filter to adjust the frequency dependence of the distortion output and a delay that is used to adjust the frequency dependence of the distortion phase. These components permit setting any particular amplitude and phase dependence of the generated distortion, but are not flexible in setting variable distortion amplitude and phase as taught in the WIPO Application. The &#39;639 patent also discloses a second path for CTB generation with adjustable amplitude and phase. 
     U.S. Pat. Nos. 6,574,389 (&#39;389 patent) and 6,593,811 (&#39;811 patent) which are incorporated by reference in their entirety herein, teach in-line distortion generation where the distortion can be generated in the main signal path with a feedback path of amplifiers altered by a non-linear element that modifies the amplifier distortion output. The in-line pre-distortion disclosed in the &#39;389 patent and &#39;811 patent provides a simple implementation to generate distortion but introduces complexity in dealing with device parasitics that affect the main signal gain and distortion generation and limit freedom in adjusting distortion generation amplitude and phase independently of main signal path gain. 
     The feedback distortion generation as shown in FIG. 2 of the &#39;811 patent can be applied to the emitter path of RF transistors, where the impedance is low and the circuit behavior may not be affected significantly by parasitic capacitance of distortion generating components such as Schottkey diodes. Some implementations can include the use of multiple Schottkey diodes and other distortion generating components (e.g., varactors) and gain adjusting elements (e.g., PIN diodes) all at once at a single emitter node. As a result the feedback networks can be built such that one transistor or gain block can simultaneously generate CSO and CTB and optionally a frequency dependence thereof. As the impedance of the feedback network changes, the gain of the transistor stage changes, which can cause an unwanted variation of the linear signal gain. By adding a variable resistor in the feedback network such as a PIN diode the feedback network impedance can be controlled to obtain or maintain a desired linear gain. One implementation of a distortion generator of this type could be used in the main path intended in the &#39;811 patent. However, main path signal levels are often high, and combining the high power capability of a main path amplifier with a capability to generate variable distortion amplitude and phase and distortion frequency dependence is complex. In another implementation feedback distortion generation in a bypass path permits independent control of distortion amplitude and phase and also frequency dependence using a small number of low power RF transistors. 
     SUMMARY 
     Systems and methods can operate to provide broadband linearization operable to generate frequency independent second and third order distortions and frequency dependent amplitude distortions. Systems can include a splitter operable to receive an input signal and to split the input signal into a first signal and a second signal, a distortion generator operable to receive the second signal and to generate a third signal, the distortion generator comprising: one or more amplifiers, one or more non-linear elements, and wherein, the one or more amplifiers and one or more non-linear elements are operable to generate second and third order distortions through adjustable bias currents, and a combiner operable to combine the first and third signals. Methods can include receiving a input signal, splitting the input signal into a first and second signal, generating one or more adjustable second and third order distortion signals from the second signal, generating one or more adjustable frequency dependent amplitude and phase distortion signals from the second signal, producing a third signal through combining the second and third order distortions signals with the frequency dependent amplitude and phase distortion signals, and combining the first and third signals. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a block diagram illustrating an example of an improved broadband linearizer. 
     
    
    
     DETAILED DESCRIPTION 
     In some implementations of this disclosure, methods, systems, and apparatuses can operate to provide a compact distortion generator with CSO and CTB output, adjustable distortion amplitude, and phase and frequency dependence with minimal impact on main signal path gain. 
       FIG. 1  is a block diagram illustrating an example of an improved broadband linearizer. In one implementation the broadband linearizer  100  can have input  105 , a signal splitter  110 , a combiner  115  and a distortion generator  120 . Following the combiner  115  there can be an output amplifier  160 , which can have additional quadrature distortion generating components as disclosed in the &#39;811 patent. Signal splitter  110  can split the input signal  105  into a main signal path  170  and a distortion path  180 . Combiner  150  recombines the distortion generator output  190  into the main signal path  170 . 
     The distortion generator  120  in some implementations can have an amplifier  122  and a balun  124 . The balun  124  can be used to generate a positive +x and negative −x phase of the input signal  180 . Each signal phase  126  and  128  can be used to drive transistor stage  130  and  140  with nonlinear feedback in the emitters. The nonlinear feedback can consist of Schottkey diodes  131 ,  132 ,  141  and  142  with individually adjustable bias current and PIN diodes  133  and  143  for impedance control. 
     Transistor  134  has gain and can generate a distortion output collector current I c1  for the positive signal input phase +x.
 
 I   c1   =A   1   x+B   1   x   2   +C   1   x   3  
 
Transistor  144  has gain and can generate a distortion output collector current I c2  negative for the negative signal input phase −x.
 
 I   c2   =−A   2   x+B   2   x   2   −C   2   x   3  
 
RF transistors can have high collector impedance representing current sources where the collector currents can be summed.
 
 I   c     —     total =( A   1   −A   2 ) x +( B   1   +B   2 ) x   2 +( C   1   −C   2 ) x   3  
 
The second order distortion can be readily generated in the summed output when A 1 =A 2 , B 1 =B 2 , and C 1 =C 2  but the third order distortion can also be generated by choosing B 1 =B 2 =0 and C 1 ≠0 or C 2 ≠0. Setting C&gt;0 generally implies that A will also become nonzero so that the linear gain becomes dependent on C 1 , however by controlling A 2  such that A 2 =A 1  there is no effect on the linear gain which can be held at zero if desired. A 1  and A 2  can readily be controlled without affecting C 1  or C 2  by adjusting current in the PIN diodes  133  and  143 .
 
     The control of the second order coefficients Band third order coefficients C can be done by adjusting the Schottkey diode currents. The input voltage to the transistor base is also present at the emitter which follows the transistor base voltage due to the high current gain of the transistors  134  and  144 . The current through the Schottkey diodes is part of the transmitter emitter current that is provided nearly 1:1 to the collectors, therefore the distortion current in the collector of a transistor can be found by analyzing the current in the Schottkey diodes. The emitter voltage of transistor  134  follows the input signalx, the current in Schottkey diode  131  can be I S1 =I o  (exp((x+V bias1 /V th )−1)) where the thermal voltage Vth=kT/q (q represents the electron charge, k the Boltzmann constant and T the temperature). I o  represents the diode saturation current. The diode can be kept in a forward bias such that the current in Schottkey diode  131  can be simplified to I s1 =I bias1  exp(x/V th ). The current for the reversed polarized Schottkey diode  135  can be represented as I s2 =I bias2  exp (−x/V th ). A Taylor series expansion of the exponential functions results in: 
               I     s   ⁢           ⁢   1       =       I     bias   ⁢           ⁢   1       (         (     x     V   th       )     +       1   /   2     ⁢       (     x     V   th       )     2       +       1   /   6     ⁢     (       (     x     V   th       )     3     )     ⁢     
     ⁢     I     s   ⁢           ⁢   2           =       I     bias   ⁢           ⁢   2       (       (       -   x       V   th       )     +       1   /   2     ⁢       (     x     V   th       )     2       -       1   /   6     ⁢     (       (     x     V   th       )     3     )                     
The sum of the emitter current for transistor  134  is the difference of the opposite currents for polarized Schottkey diodes  134  and  135  and can be represented by.
 
     
       
         
           
             
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     Second order distortion content can be created by adjusting the bias currents of Schottkey diodes  131  and  132  to 
               B   ⁢           ⁢   1     =       (       I     bias   ⁢           ⁢   1       -     I     bias   ⁢           ⁢   2         )     *     1   2     *       (     1     V   th       )     2             
where the two bias currents are not equal. The second order distortion component can be minimized when the bias currents of Schottkey diodes  131  and  132  are equal and can result in a third order component of
 
               C   1     =       (       I     bias   ⁢           ⁢   1       +     I     bias   ⁢           ⁢   2         )     *     1   2     *     1   3     *         (     1     V   th       )     3     .             
The linear signal term can be
 
     
       
         
           
             
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     At each transistor  134  and  144  there can be a PIN diode  133  and  143  connected to the emitter of each transistor respectively. At transistor  144  the Schottkey diodes  141  and  142  can each have a bias current equal to zero where a non-zero bias current on PIN diode  143  can result in a impedance of R PIN =Z nom *I nom /I bias     —     pin  where Z nom  represents the PIN diode  143  nominal impedance at a reference bias current I nom  and I bias     —     pin  is the actual PIN diode  143  bias current. A signal −x at the base of transistor  144  can result in an emitter current of I e2 =−x/R PIN =−x*I bias     —     pin /(Z nom *I nom ) and can result in A 2 =I bias     —     pin /(Z nom *I nom ). 
     PIN diode  143  can be a linear element and can result in of distortion coefficients B 2  and C 2  approximating zero if the currents in Schottkey diodes  141  and  142  at transistor  144  are set to zero. The combined collector current of transistor  134  and  144  can be C 1 *x 3  in the example given and can result in the generation of a third order distortion without second order distortion or linear gain. By adjusting the Schottkey diode  131 ,  132 ,  141  and  142  and PIN diode  133  and  143  currents, the linear gain, second and third order distortion components can be varied in amplitude including the sign. 
     A multitude of transistor pairs  151  and  152  can be driven with additional distortion generating elements in their emitters with Schottkey diodes  153  and  154 . The collector outputs of transistors  151  and  152  can be provided to one or more filters  155  and can provide frequency dependent amplitude and phase distortion. The outputs of the one or more filters  155  can be connected to distortion path  180  and can provide a number of frequency dependent amplitude and phase distortion profiles where a combination thereof can be set by adjusting the various Schottkey diode bias currents. For instance a distortion profile with increasing distortion as a function of frequency can be set with an in-phase distortion vector. The degree of frequency dependence can be adjusted by adjusting the amount of distortion generated by frequency dependent  150  and frequency independent  130  and  140  stages. Optionally variable attenuators  121  can be added at distortion generator input and output (not shown) to further add to the level control. 
     The distortion generator output signal  190  can be combined with the main path signal  170 ; adjustment of the distortion phase relative to the main path signal phase can be done by setting the correct delay on the main path signal  170 , typically to match the delay of the distortion generator  120 . The distortion generator  120  can be used to cancel distortion of a laser diode in an optical transmitter. The distortion phase of the CTB generated by the laser can be related to that of one or more components of the CSO generated by the laser, they are just part of the same nonlinear transfer curve. Therefore a combination of CSO and CTB can be generated by the same transistors and a separate path for CTB as in the WIPO Application is not required. 
     In other implementations a plurality of distortion generators (not shown) can be placed in the distortion generating bypass path  190  each with a different amplitude and phase dependence on frequency.