Abstract:
A test system automatically locates impedance mismatched, energy reflection discontinuities (e.g., bridged taps) along a wireline communication link, by coupling a linearly stepped sinusoidal waveform to a measurement location of the wireline link. The applied waveform propagates down the link and is reflected back from the energy reflection discontinuities. A line monitoring receiver coupled to the measurement location samples the response of the line to the swept waveform. A response processor executes a frequency domain reflectometry algorithm to analyze the frequency response of the wireline to the stepped frequency waveform. It then generates an output representative of distances from the measurement location to the bridged taps.

Description:
FIELD OF THE INVENTION 
     The present invention is directed to communication systems, and is particularly directed to a system for automatically locating and thereby facilitating removal of energy reflection anomalies, such as bridged taps, and the like, that may impair digital communications along a wireline telecommunication link. 
     BACKGROUND OF THE INVENTION 
     In the face of the increasing demand for a variety of to digital communication services (such as, but not limited to internet services), telecommunication service providers are continually seeking ways to optimize the bandwidth and digital signal transport distance of their very substantial existing copper plant, that was originally installed for the purpose of carrying nothing more than conventional analog (plain old telephone service or POTS) signals. 
     In addition to the inherent bandwidth limitations of the (twisted pair) copper wire medium, service providers must deal with the fact that in-place metallic cable plants, such as that shown at  10  in the reduced complexity network diagram of FIG. 1, linking a central office  12  with a subscriber site  14 , typically contain one or more anomalies, such as but not limited to load coils (used to enhance the wireline&#39;s three to four kilohertz voice response), and bridged taps  16 , to which unterminated (and therefore reflective) lateral twisted pairs  18  of varying lengths may be connected. 
     Because these discontinuities cause a portion of the energy propagating along the wireline link to be reflected back in the direction of the source, at the high frequencies used for digital data communications (e.g., on the order of one MHz), such reflections can cause a significant reduction in signal amplitude, when (counterphase) combined with the original signal, thereby disrupting digital data service. In order to locate these reflection points, it has been conventional practice to employ interactive, time domain reflectometry (TDR), which relies upon the ability of a skilled technician to make a visual interpretation of a displayed TDR waveform, and thereby hopefully identify the bridged taps, and the lengths of any laterals that may extend therefrom. Because this process is subjective, it is not only imprecise, but is very difficult to automate. 
     SUMMARY OF THE INVENTION 
     In accordance with the present invention, shortcomings of conventional TDR-based schemes for locating energy reflecting anomalies, such as bridged taps and the like, along a wireline telecommunication link, are effectively obviated by means of an objective, frequency domain reflectometry (FDR)-based mechanism. The test mechanism of the invention may be implemented in a processor-controlled test head installed in a central office, or as part of test signal generation and processing circuitry of a portable craftsperson&#39;s test set. 
     The test head contains test signal generation and processing circuitry, that is operative to execute a frequency domain reflectometry (FDR) algorithm, through which the line is stimulated by means of a linearly swept (stepped) sinusoidal waveform, to invoke a line response that is readily measured and analyzed to reveal the locations (distance from the source) of any impedance mismatch reflection discontinuities (e.g., bridged taps and the like). 
     A control processor is programmed to generate a sinusoidal waveform that is linearly stepped from a minimum frequency of 0 Hz (DC) to a maximum frequency f max . As the frequency of the test waveform applied to the line is swept, the signal level at a line access point is monitored via an input amplifier, digitized, and then and stored in a signal measurement buffer. The amplitude of the measured signal response will exhibit a variation with frequency that is a composite of fluctuations in impedance due to any reflection points (e.g., bridged taps) along the line. 
     Once captured, the response data is weighted to optimize the accuracy of the analysis. Any DC level is removed, and a window, such as a Hanning window, is used to remove discontinuities between start and end values of the captured data set, to avoid spurious results. A loss compensation function is applied to the modified data set, to compensate the frequency response characteristic of the line for loss over distance and frequency. A frequency-dependent propagation constant is derived in terms of the resistance, inductance, capacitance and conductance of the line per unit length. The real part of the propagation constant is the attenuation along the line per unit length. The attenuation of the envelope of a signal propagating along the line is an exponential function of the propagation constant. The effect on the frequency response waveform is that amplitude decay is less pronounced for reflected signals propagating on shorter loops, since the shorter distance offsets the effects of the loss at high frequencies. Because the length of the line under test is unknown, a compromise between the two extremes provides compensation for the overall frequency response waveform irrespective of distance. 
     The loss-compensated data is then processed by a frequency analysis operator, such as discrete Fourier transform (DFT), which decomposes the composite line signal response into frequency bins associated with the individual reflectors&#39; frequency fluctuations. A threshold is established for the contents of the frequency bin data produced by the DFT, in order to distinguish between useful or significant energy and spurious energy. A frequency bin is considered to contain significant energy, if its contents exceed the threshold for that bin number. Any frequency bin whose contents exceed its threshold are subjected to frequency domain reflectometry (FDR) analysis. 
     In the context of detecting bridge taps along a wireline telecommunication link, a waveform propagating downstream along the wireline combines with a waveform reflected from a bridged tap and returning upstream along that wireline. Since the downstream and upstream propagating waveform components have the same frequency, the composite waveform will have a local minimum due to destructive interference at some time delay when the arguments of the two waveform components differ by π radians. Nulls will occur for other frequencies, where the arguments of the waveform components differ by odd multiples of π. A linear sweep of a wireline having a single reflection point (e.g., bridged tap) will produce nulls at frequencies f o , 3f o , 5f o , 7 o , etc. In general, the null repetition rate in the frequency domain F n  may be given by: F n =1/2f n , where f n  is the lowest frequency at which a null occurs when the delay t=t n . F o  corresponds to t o  and, in general, F n  corresponds to t n  and is the same as the round-trip delay of the signal from the line access location to the point of reflection along the line and back. 
     In order to determine the length of time required for the waveform to propagate to the impedance-mismatch reflection point, it may be observed that to is representative of the total time required for the downstream propagating waveform to be reflected back to the access location at which the measurement is taken. This one-way delay t i  is equal to t o . To determine the distance of this reflection point from the access location, the propagation velocity v p  of the waveform along the wireline, which is readily calculated, is employed. The distance from the access location to the location of the impedance mismatch reflection is inversely proportional to frequency, and the minimum resolvable distance D min =v p /2f max . 
     Where the line under test contains plural discontinuities, the response waveform seen at the signal measurement point will contain multiple components produced by the plurality of reflection points. Since these reflected waveforms components are generally associated with impedance discontinuities caused by physical characteristics in the wireline separated by varying distances from the source, the delays associated with these reflections will be mutually different, so that their frequencies will be mutually different. As each delay produces its own unique frequency, then by identifying the various frequencies, the two-way delay times of a reflection from a wireline discontinuity may be readily determined, so that the distance to the impedance discontinuity may be determined. 
     To determine the individual values of two-way delay time, the frequency response waveform produced by stimulating the wireline under test with a linearly swept sinusoidal waveform is sampled at discrete frequency steps of (f max /N). The DFT produced will yield values that area proportional to the magnitudes of the various null repetition rates F k . The contents of the first frequency bin are the DC component of the swept response, while a respective bin m contains the magnitude of the null repetition rate (m−1)F o , for m=2,3,4, . . . N/2. Namely, the various energy bins of the response represent energy associated with the time delays t o , 2t o , 3t o , etc., and contain the magnitude of the waveforms delayed by (m−1) t o  for m=2,3,4, . . . N/2. 
     As a result, the bins of the DFT, which represent different round trip delay times of the swept waveform, can be employed to determine the distances from the access location to energy-reflecting anomalies. The distance to a reflection point may be determined by multiplying the one-way delay by the velocity of propagation of the swept waveform. In general, the bins of the response represent distances that are integral multiples of the delay t o . Namely, the distance D m−1  associated with a bin m−1 is equal to (m−1)t o v p /2 or [(m−1)t o ]v p /2 for m=2,3,4, . . . N/2. Thus, there is a one-to-one correspondence between the bins of DFT and distances to the reflection points along the wireline. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a reduced complexity network diagram showing an unterminated lateral twisted pair extending from a bridged tap installed in a wireline communication link between a central office and a subscriber circuit; 
     FIG. 2 diagrammatically illustrates a reduced complexity embodiment of an automated bridged tap detection arrangement in accordance with the present invention; 
     FIG. 3 is a flow chart showing the steps of the automated wireline reflection anomaly detection mechanism of the present invention; and 
     FIG. 4 is a frequency sweep response characteristic obtained in the course of executing the mechanism of FIG. 3 showing periodic nulls at frequencies f o , 3f o , 5f o , 7 o , etc. 
    
    
     DETAILED DESCRIPTION 
     Before detailing the automated (FDR)-based energy reflection location mechanism of the present invention, it should be observed that the invention resides primarily in an arrangement of conventional communication hardware components and attendant supervisory communications microprocessor circuitry and application software therefor, that controls the operations of such components and analysis of signal waveforms interfaced therewith. In a practical implementation that facilitates their incorporation into telecommunication link test equipment (such as that which may be installed at a central office or resident in a portable test device), the inventive arrangement may be readily implemented using a general purpose digital computer, or field programmable gate array (FPGA)-configured, application specific integrated circuit (ASIC) chip sets. In terms of a practical hardware implementation of such chip sets, digital ASICs are preferred. 
     Consequently, the configuration of such components and the manner in which they may be interfaced with a (copper) wireline communication link have, for the most part, been illustrated in the drawings by readily understandable block diagrams and flow charts, which show only those specific details that are pertinent to the present invention, so as not to obscure the disclosure with details which will be readily apparent to those skilled in the art having the benefit of the description herein. Thus, the block diagram and flow chart illustrations of the Figures are primarily intended to show the major components and functional modules of the system of the invention in convenient functional groupings, whereby the present invention may be more readily understood. 
     Attention is now directed to FIG. 2, wherein a reduced complexity embodiment of an automated bridged tap detection arrangement in accordance with the present invention is shown diagrammatically as comprising a processor-controlled test head  20 , such as may be installed in a central office, or as part of test signal generation and processing circuitry of a portable craftsperson&#39;s test set. The test head circuitry  20  is adapted to be interfaced at an access location  21  with a line under test (LUT)  22  (corresponding to the metallic twisted pair  10  of FIG. 1, described above), by means of a line-driver amplifier  24  and an input/receiver amplifier  26 . The line-driver amplifier  24  is coupled to access location  21  of the metallic line pair  22  through a pair of source resistors  27 ,  28 , each of which has an impedance equal to one-half the impedance (Zo) of the line  22 . 
     Pursuant to the invention, the test head  20  contains test signal generation and processing circuitry, that is operative to execute a frequency domain reflectometry (FDR) algorithm, through which the line  22  is stimulated to produce a response that is readily measured and analyzed to reveal where along the line reflection points are located. To this end, the test head contains a test control digital processor  30  that sources digitally created test signals in the form of a frequency-swept sinusoidal waveform. In addition, it conducts a digitally-based analysis of the line&#39;s response to the those test signals in terms of fluctuations in impedance caused by the various reflections. Since the test signals are in the form of a frequency swept analog tone, the control processor is interfaced with the line-driver and input receiver amplifiers by means of associated digital-to-analog and analog-to-digital converter circuits. 
     More particularly, referring now to the overall functional sequence of the automated reflection anomaly detection mechanism of the present invention shown in the flow diagram of FIG. 3, at an initial step  301  the test head&#39;s processor generates a varying frequency sinusoidal waveform, that is converted into an analog sinusoidal tone signal and applied to the line-driver amplifier  24 , which drives the line under test. In accordance with a preferred embodiment the frequency of the waveform is varied in a linear, stepwise manner, for example beginning at minimum frequency such as 0 Hz and stepping in finite incremental frequency steps up to a maximum frequency. Conversely, the frequency variation may begin at an upper frequency and proceed to a minimum frequency, without a loss in generality. As the frequency of the sinusoidal waveform is swept, the signal level at the test access point  21  is monitored via the input amplifier  26 , digitized, and then and stored in a signal measurement buffer (not shown) in step  302 . The amplitude of the measured signal response will exhibit a variation with frequency that is a composite of the fluctuations in impedance due to any reflection points along the LUT. 
     In step  303 , the stored response data is modified to optimize the accuracy of the analysis. In particular, as shown at step  303 - 1 , any DC level in the stored data is removed. Secondly, as shown at step  303 - 2 , a code filtering (e.g., Hanning) window is applied to the data. Steps  303 - 1  and  303 - 2  serve to remove discontinuities between the start and end sample values of the captured data set, that might otherwise cause spurious results. 
     Next, in step  304 , a loss compensation function (LCF) is applied to the adjusted data set, in order to compensate the frequency response characteristic of the line for loss over distance and frequency. In particular, the line under test can be characterized in terms of its resistance (R), inductance (L), capacitance (C), and conductance (G) parameters per unit length, which are available from tabulated industry-available sources for the type of wire. From these parameters, a frequency-dependent propagation constant T can be derived as: 
     
       
         τ=α+ j β=(( R+jwL )( G+jwC )) 1/2 , where w=2 πf.   
       
     
     The real part of the propagation constant, α(f), is the attenuation along the line per unit length. Since the envelope of a signal propagating along the line as a function of distance is attenuated by e −α(f)t , α(f) can be determined. 
     The effect on the frequency response waveform is that amplitude decay is less pronounced for reflected signals propagating on shorter loops, since the shorter distance offsets the effects of the loss at high frequencies, due to the effects of α(f). Since the actual length of the line under test is unknown, a compromise between the two extremes is employed, to provide some amount of compensation for the overall frequency response waveform for all distances of interest. 
     In order to determine the coefficient of the exponential attenuation function in terms of frequency, it is necessary to reduce the number of degrees of freedom of the total loss function. As the maximum frequency of the swept sinusoidal waveform is known, a priori, a loss compensation function based upon the mid frequency point of the sweep f mid =f max /2 may be employed. As will be described in detail below with reference to null vs. frequency response diagram of FIG. 4, from this mid frequency, f mid , a corresponding resolution distance d mid  is defined as: 
     
       
           d   mid   =V */4 f   mid   
       
     
     An ‘average loss’ value η can be derived as: 
     
       
         η= e   −α(fmid)dmid . 
       
     
     The loss compensation function LCF can therefore be defined as: 
     
       
           LCF=exp ((−2 ln (η)/ f   max ) f ). 
       
     
     The loss-compensated data is then processed in step  305  by means of a frequency analysis operator, such as discrete Fourier transform (DFT), which decomposes the composite line signal response into frequency bins associated with the individual reflectors&#39; frequency fluctuations. In step  306 , a threshold is established for the contents of the frequency bin data produced by the DFT, in order to distinguish between significant (useful) and spurious energy. The threshold employed is defined as: 
     
       
         Threshold (bin no.)=[(StartValue−EndValue)*exp(−bin no.*slope)]+EndValue. 
       
     
     The parameters StartValue, EndValue and slope are dependent upon the test head circuitry&#39;s gain and swept bandwidth, and are readily empirically determined. A frequency bin is considered to contain significant energy, if its contents exceed the threshold for that bin number. In step  307 , any frequency bin whose contents exceed its threshold are subjected to waveform analysis of the type used in frequency domain reflectometry. 
     More particularly, for an arbitrary waveform v(t) that is the sum of two waveforms of some frequency f o , a minimum will occur in v(t) at some delay t o  of one waveform relative to the other. For the case of a wireline cable plant, this occurs when a waveform v o  propagating downstream along the wireline is combined with a waveform v 1  reflected from an anomaly, such as a bridged tap and returning upstream along that wireline. In general, the combination of these two waveforms can be expressed as: 
     
       
           v ( t )= v   o ( t )+ v   1 ([ t−t   o ]) 
       
     
     
       
           v ( t )= v   o  sin (2 πf   o   [t−t   o ]). 
       
     
     Since the downstream and upstream propagating waveform components have the same frequency, v(t) will have a local minimum due to destructive interference at some time delay to when the arguments of v o  and v 1  differ by π radians. Namely, 
     
       
         (2 πf   o   t )−(2 πf   o   [t−t   o ])=π. 
       
     
     Dividing this expression by 2πf o t and solving for t o , yields: 
     t o =1/2f o =T o /2, where the period T o  of the waveform is 1/2f o . 
     As shown in FIG. 4, where t=t o , nulls in v(t) will occur for other frequencies f k , where f k &gt;f o , and the arguments of v o  and v 1  differ by odd multiples of π. If k is a positive integer, the nulls will occur when: 
     
       
         (2 πf   k   t )−(2 f   k   [t−t   o ])=2 πf   k   t   o =(2 k +1)π. 
       
     
     Letting the period T k =1/f k , then 
     
       
         2 πf   k   t   o =2 πt   o   /T   k =2π( T   o /2)/ T   k =(2 k +1)π. 
       
     
     
       
           T   o   /T   k =(2 k +1) 
       
     
     Substituting T o =1/f o , T k =1/f k : 
     
       
           f   k   /f   o =(2 k +1), or 
       
     
     
       
           f   k   =f   o (2 k +1), for  k= 0, 1, 2, . . .  
       
     
     The periodicity of the nulls can be seen by examining the difference in frequency between two adjacent nulls f m  and f m+1 . From the foregoing, f m+1 −f m =f o (2[m+1]+1)−f o  (2m+1)=2f o , for m=0, 1, 2, . . . 
     This means that conducting a linear sweep of a wireline having a single reflection point (e.g., bridged tap) will produce nulls in the frequency response at frequencies f o , 3f o , 5f o , 7 o , etc., as shown in FIG.  4 . 
     Denoting F o  as the repetition rate of the nulls for t=t o  in the frequency domain, then: 
     
       
           F   o =1/(period of the null)=1/( f   m+1   −f   m )=1/2 f   o π. 
       
     
     In general, the null repetition rate in the frequency domain F n  is given by: F n =1/2f n , where f n  is the lowest frequency at which a null occurs when the delay t=t n . 
     From the above relationships, F o  corresponds to t o  and, in general F n  corresponds to t n , and is the same as the round-trip delay of the signal from the source or access location to the point of reflection along the line and back. In order to determine the length of time required for the waveform to propagate to the impedance-mismatch reflection point, it may be observed that t o  is representative of the total time required for the downstream propagating waveform to be reflected back to the access location  21  at which the measurement is taken. This one-way delay t i =t o . To determine the distance of this reflection point from the access location, the propagation velocity v p  of the waveform along the wireline must be known. In general, if ∈ r  is the dielectric constant of the wireline insulation, c is the velocity of light in free space, and μ r  is relative permeability, then the propagation velocity in the wireline may be expressed as: v p =C(∈ r μ r ) −1/2 . Knowing the type of cable from industry available specifications allows the propagation velocity (typically on the order of ⅔ the velocity of light) to be readily determined. 
     The distance D from the access location to the location of the impedance mismatch reflection is given by the expression: 
     
       
           D=t   i   v   p   =v   p   t   o /2 =T   o   v   p /4. 
       
     
     Thus D is proportional to T o /4, which the one-quarter wavelength of the sinusoid waveform having a frequency f o . Substituting T o =1/f o , yields D=v p /4f o . Namely, the distance D is inversely proportional to frequency. This means that the minimum resolvable distance D min =v p /2f max . 
     As pointed out above, the response waveform v(t) seen at the signal measurement point will contain components produced by a plurality of reflection points as: 
     
       
           v ( t )= v   o ( t )+ v   1 ( t−t   o )+ v   2 ( t−t   1 )+ v   3 ( t−t   2 )+ . . .  v   n ( t−t   n−1 ). 
       
     
     Since these reflected waveforms are generally associated with impedance discontinuities caused by physical characteristics in the wireline separated by varying distances from the source, the delays t o , t 1 , . . . t n−1 , associated with these reflections will be mutually different, so that the values T o /2, T 1 /2, . . . T n−1 /2 and thus the frequencies f o , f 1 , . . . f n−1  will be mutually different. Since f n  is unique for each delay, then by identifying the various f n , the two-way delay times t n  of a reflection from a wireline discontinuity may be readily determined. As demonstrated above, once the time delay is known, the distance D to the impedance mismatch discontinuity (e.g., bridged tap) may be readily determined. 
     To determine the individual values of two-way delay time t n , the frequency response waveform a(f) produced in response to stimulating the wireline under test with a linearly swept sinusoidal waveform is sampled at discrete frequency steps of (f max /N), resulting in the sampled waveform a(k) being stored in step  302 , as described above. For a radix-two buffer size of N points, the output of the DFT operation of step  305  will yield values that are proportional to the magnitudes of the various null repetition rates F k . If the maximum frequency f max  of the swept sinusoid waveform is 2f o , the minimum resolution of the DFT is: 
     
       
         1 /f   max =1/2 f   o   =F   o   =t   o  (seconds). 
       
     
     Denoting the contents of frequency bin m as A(m) of the DFT of a(k), then the contents A(l) of the first frequency bin are the DC component of the swept response, while the bin m contains the magnitude of the null repetition rate (m−1)F o , for m=2,3,4, . . . N/2. 
     Namely, the various energy bins of the response A represent the energy in a(f) associated with the time delays t o , 2t o , 3t o , etc., and A(m) contains the magnitude of the waveforms delayed by (m−1)t o  for m=2,3,4, . . . N/2. 
     Thus, the bins of A, which represent different round trip delay times of the swept waveform, can be employed to determine the distances from the access location to the energy-reflecting anomalies. The distance D RP  to a respective reflection point RP′ is readily determined by multiplying the one-way delay t RP  by the velocity of propagation v p  of the waveform. 
     In general the bins of the response A represent distances that are integral multiples of the delay t o . Namely, D m−1 =(m−1)t o v p /2=[(m−1)t o ]v p /2 for m=2,3,4, . . . N/2, so that there is a one-to-one correspondence between the bins of DFT and the distances to the reflection points along the wireline. 
     EXAMPLE 
     Using the measurement arrangement diagrammatically illustrated in FIG. 2, a twisted pair wireline cable was stimulated with a sinusoidal waveform, the frequency of which was linearly varied in a stepwise manner from f min =0 Hz to f max =2 MHz. The source output impedance Zo was 100; namely, each source resistor  27  and  28  had an impedance of Zo/2 or fifty ohms. To accommodate a buffer size of 512 addresses, the frequency of the signal source was stepped at increments of 2 MHz/512 or 3906.25 Hz per step. At each frequency step, the line was allowed to settle and the steady state response was sampled and stored for that frequency bin. The propagation velocity v p  for the cable type was 634×106 ft/sec. 
     For N=512, the output of the DFT for the stored response will produce 256 bins. As pointed out above, the first bin contains the DC component and is of no interest. As a non-limiting example, the threshold for the m=40th bin was exceeded, indicating the presence of significant energy in that bin. Thus, a reflection occurred after a time delay of (m−1)/2f max =(40−1)/(2×2000000) or approximately 9.75 μsec from the source location  21 . Using the above expression for distance, this means that the distance from the source to the impedance discontinuity associated with energy for bin number  40  is 9.75 μsec×v p =9.75×634000000/2 ft/sec=3090 ft. 
     As will be appreciated from the foregoing description, the shortcomings of conventional TDR-based schemes for locating energy reflecting anomalies, such as bridged taps and the like, along a wireline telecommunication link, are effectively by using frequency domain reflectometry-based mechanism of the present invention. By stimulating the line with a linearly stepped frequency sinusoidal waveform, the test head evokes a composite waveform response which, when subjected to frequency domain reflectometry analysis, yields distance data representative of locations of the energy reflection discontinuities. Service personnel may then determine what action, if any, needs to be taken relative to the located anomaly. At least knowing the location of any discontinuity allows for a more accurate determination of whether a POTS line is qualified for digital service, or requires action by a field technician to remove one or more discontinuities. 
     As pointed out above, the frequency bins of the response waveform represent distances that are integral multiples of delay, so that there is a one-to-one correspondence between the bins of discrete Fourier transform and distances to the reflection points along the wireline. Since the highest or maximum frequency of the stepped sweep is inversely proportional to the time resolution of the reflection points, the higher the maximum frequency, then the shorter the time resolution. In terms of distance, a higher maximum frequency facilitates resolving shorter (closer) discontinuities. Increasing the number of frequency steps enables longer propagation times to be measured, and thereby allows farther away discontinuities to be located. 
     While we have shown and described an embodiment in accordance with the present invention, it is to be understood that the same is not limited thereto but is susceptible to numerous changes and modifications as known to a person skilled in the art, and we therefore do not wish to be limited to the details shown and described herein, but intend to cover all such changes and modifications as are obvious to one of ordinary skill in the art.