Abstract:
A method and apparatus for digital quadrature lock-in detection capable of receiving magnetic resonance signals or electron spin resonance signals at high sensitivity. The method starts with digitizing a signal wave consisting of a magnetic resonance or electron spin resonance signal. The digitized signal wave is multiplied by digitized reference waves of sine and cosine functions to obtain signals of real and imaginary parts which are 90° out of phase (multiplying step). The frequencies of the sine and cosine functions are varied according to the observation width. The multiplying step is repeated.

Description:
BACKGROUND OF THE INVENTION  
       [0001]     1. Field of the Invention  
         [0002]     The present invention relates to a method and apparatus for digital quadrature lock-in detection in magnetic resonance.  
         [0003]     2. Description of Related Art  
         [0004]     (Analog Quadrature Detection)  
         [0005]      FIG. 7  is a block diagram showing an example of the configuration of an NMR (nuclear magnetic resonance) spectrometer. The spectrometer has a pulse sequencer  1  producing RF (Radio Frequency) pulses that enter a direct digital synthesizer (DDS) daughter card  2 , where given processing is performed. The daughter card  2  consists of a digital direct synthesizer (DDS)  2   a , a D/A (digital to analog) converter (DAC)  2   b , and a low-pass filter  2   c . The DDS daughter card  2  produces pulses of 9 to 12 MHz.  
         [0006]     The RF pulse then enters a following transmitter  3 . The transmitter  3  includes a mixer  3   a  and an attenuator (ATT)  3   c  for attenuating the output from the mixer  3   a , which mixes the output from the DDS daughter card  2  and the output from a distributor  3   b . The output from the attenuator  3   c  is amplified by a following power amplifier  4  and applied to a probe  6 . An NMR signal detected by the probe  6  is passed to an OBS receiver  10  via a duplexer (DPLX)  5  and a preamplifier  7 .  
         [0007]     In the OBS receiver  10 , the NMR signal is mixed with a locally generated (LO) signal from a frequency synthesizer (FSY)  20  by an image reject mixer (IRM)  11 . The signal from the synthesizer  20  is selected by an FSY selector  12  and applied to the image reject mixer  11 , where the signal is mixed with the signal from the preamplifier  7 . The output signal from the mixer  11  has an intermediate frequency (IF). The output from the image reject mixer  11  is passed through a bandpass filter  13  that passes only frequencies of 9 to 12 MHz. The output signal from the filter  13  is split into two by a splitter  14 .  
         [0008]     An intermediate frequency (IF) is generated by a digital direct synthesizer (DDS)  15  and passed to mixers  17   a  and  17   b  through D/A (Digital to Analog) converters (DACs)  16   a  and  16   b , respectively. The two signals from the splitter  14  are mixed with intermediate-frequency (IF) signals which are 90° out of phase at the mixers  17   a  and  17   b , respectively. The output signals from the mixers  17   a  and  17   b  are passed through AF (audio frequency) filters  18   a  and  18   b , respectively, to remove unwanted RF components. The output signals from the filters  18   a  and  18   b  are converted into digital signals by A/D (Analog to digital) converters (ADCs)  31   a  and  31   b , respectively. The resulting data are temporarily held in a signal processing portion  32 . One of the two digital signals is herein referred to as a real part, while the other is referred to as an imaginary part. In this case, any one of them may be a real part.  
         [0009]     (Digital Quadrature Detection)  
         [0010]      FIG. 8  is a block diagram showing a second example of the configuration of an NMR spectrometer. Like components are indicated by like reference numerals in both  FIGS. 7 and 8 . In analog quadrature detection shown in  FIG. 7 , two signals, which originate from the digital direct synthesizer (DDS)  15  and are applied as IF signals to the mixers, are out of phase. In contrast, in the configuration shown in  FIG. 8 , one phase signal is converted into an analog signal by a DAC  16   b  and then mixed with an NMR signal at a mixer  17   b . However, when the signal is accepted into the ADC  31   b , it is necessary that the signal be oversampled at a frequency more than twice as high as the required spectral width SW.  
         [0011]      FIG. 1  is a conceptual diagram illustrating digital quadrature detection underlying the concept of the present invention, and depicts subsequent processing on digital data obtained by the signal processing portion  32 . Time domain data stored in a data buffer  40  are copied as two data sets. One data set is multiplied by sin Ωt, while the other is multiplied by cos Ωt. The products are passed through digital low-pass filters (D-LPFs)  42   a  and  42   b , respectively, and stored in data buffers  2 R and  2 I, respectively. The stored data sets are taken as real and imaginary data, respectively. The data buffers  2 R and  2 I are indicated by  43   a  and  43   b , respectively.  
         [0012]     (Summary and Principle of Lock-In Amplifier)  
         [0013]     Summary and principle of a lock-in amplifier are next described.  FIG. 2  illustrates lock-in detection and the principle of a lock-in amplifier. It is assumed that an observed signal consists of a single frequency. The observed signal can be given by sin(ωt+α). A case is discussed in which a reference signal is prepared and brought into coincidence with the observed signal in terms of frequency or the frequency is swept for coincidence.  
         [0014]     If the coincidence is achieved, the reference signal can be given by sin(ωt+β). If the product of the reference signal sin(ωt+β) and observed signal sin(ωt+α) is calculated by a multiplier  45 , formulas of products and sums of trigonometric functions state that the output from the multiplier  45  is given by 
 
cos(2ωt+α+β)−cos(α−β) 
 
 Of this formula, the former term is an AC component varying with time. The latter term is a DC component not varying with time. 
 
         [0015]     Only the DC component −cos(α−β) can be obtained by cutting off RF components by a low-pass filter (LPF)  46  having a cutoff frequency lower than 2ω and passing only low-frequency components. Generally, noise can be blocked by cutting off the region other than a band of interest by a filter. In the case of this lock-in amplifier, noise components located outside a band of reference signal ±about cutoff frequency can be blocked because the cutoff frequency of the low-pass filter  46  can be set to a very low frequency. As a result, low-noise detection is enabled. Hence, this lock-in amplifier is often used for high-sensitivity measurements. This detection system is referred to as the lock-in amplifier.  
         [0016]     (Quadrature Lock-In Detection)  
         [0017]     Quadrature lock-in detection is next described. The reference signal shown in  FIG. 2  is sin(ωt+β). The phase is not definite. The signal cannot be observed depending on the value of β and thus the phase of the observed signal cannot be known. (Usually, the phase is to be optimized.) To avoid this, a quadrature lock-in amplifier is available.  
         [0018]      FIG. 3  illustrates quadrature lock-in detection. Like components are indicated by like reference numerals in both  FIGS. 2 and 3 . In this quadrature lock-in amplifier, an observed signal is multiplied by reference signals by means of multipliers  45  and  45   a , respectively. The observed signal is split into two parts which are multiplied by reference signals which are 90° out of phase (e.g., sine wave and cosine wave, respectively). The products are passed through low-pass filters  46  and  46   a , respectively, in the same way as in the above-described example. They are observed as two out of phase signals. Where the reference signal is sin(ωt+β), the DC output signal from the low-pass filter  46  is −cos(α−β). Where the reference signal is cos(ωt+β), the DC output signal from the low-pass filter  46   a  is sin(α−β).  
         [0019]     Thus, signals can be received with at least one of the two channels without depending on the phases of the observed signal α and reference signal β. Furthermore, the phase of the observed signal relative to the reference signal can be known. Note that symbols and coefficient 1/2 are omitted in  FIGS. 2 and 3 .  
         [0020]     (Digital Lock-In Detection)  
         [0021]     Digital lock-in detection is next described. The above-described lock-in amplifier can be digitized.  FIG. 4  illustrates digital lock-in detection. An analog observed signal sin(ωt+α) is entered. An A/D converter  47  converts the observed signal into a sequence of digital numerical values varying with time. The converted observed signal is input to one input terminal of a multiplier  48 . A digital reference signal sin(ωt+α) is input to the other input terminal of the multiplier  48 .  
         [0022]     The reference signal is multiplied by a sequence of numerical values corresponding to a sine wave. The obtained sequence of numerical values is passed through a digital low-pass filter  49 . In this way, a sequence of digital numerical values −cos(α−β) corresponding to a DC signal can be obtained. A lock-in amplifier that has been digitized in this way can also be available. In addition, a digital quadrature lock-in amplifier can also be achieved by combination with the circuit shown in  FIG. 3 .  
         [0023]     An apparatus of this kind is known as disclosed, for example, in Japanese Patent Laid-Open No. H10-99293 (paragraphs 0008-0012; FIGS. 1 and 2). In this technique, real and imaginary part data are obtained by DPSD complex detection based on a 4-fold oversampling technique, and the difference in amplitude between the real and imaginary part data is reduced. Furthermore, a technique regarding improvement of a quadrature phase detection technique for performing quadrature phase detection of a signal from the receiver coil of a magnetic resonance imaging apparatus is known (see, for example, Japanese Patent Laid-Open No. H5-317285 (paragraphs 0016-0034; FIGS. 1 and 3).  
         [0024]     Another technique for reconstructing a magnetic resonance signal is also known as disclosed, for example, in Japanese Patent Laid-Open No. H4-357937 (paragraphs 0015-0022; FIG. 6). The magnetic resonance signal is separated into two signals by detection using two reference waves having resonance frequencies that are 90° out of phase. The detected signals are Fourier transformed and corrected in the Fourier space, thus removing noise. Then, the signal is reconstructed.  
         [0025]     In the case of the aforementioned analog quadrature detection, two different A/D converters (ADCs) are used. Therefore, it is difficult to make uniform the converters in gain and DC offset. Accordingly, there is the problem that image artifacts and center glitch (artifacts caused by frequency offset) tend to be produced. Furthermore, generally, in magnetic resonance phenomena (especially, nuclear magnetic resonance phenomena such as NMR and MRI), quite weak energies are treated. Therefore, the sensitivity is low in principle. Accordingly, improving the sensitivity is quite important in developing magnetic resonance apparatus.  
       SUMMARY OF THE INVENTION  
       [0026]     It is an object of the present invention to provide a method and apparatus for digital quadrature lock-in detection permitting high-sensitivity observation of signals in magnetic resonance spectroscopy.  
         [0027]     A first aspect of the present invention provides a method of receiving a magnetic resonance signal or electron spin resonance signal by a digital quadrature detection technique. A digitized signal wave is multiplied by digitized reference waves of sine and cosine functions, resulting in signals of real and imaginary parts which are 90° out of phase. The frequencies of the sine and cosine functions are varied according to the observation width. This calculation in which the signal wave is multiplied by the reference waves is repeated.  
         [0028]     A second aspect of the present invention provides an apparatus for receiving a magnetic resonance signal or electron spin resonance signal by a digital quadrature detection technique. The signal is sampled at a frequency higher than a required frequency band, producing digital data. The digital data are detected by a quadrature digital detection technique. At this time, the signal wave is repetitively multiplied by plural different frequency-variable sine or cosine functions between {a fixed frequency−(required bandwidth)/2)} and {the fixed frequency+(required bandwidth)/2)}, using the same principle as a digital lock-in amplifier. The resulting signals are passed through a narrow-band digital low-pass filter to cut off unwanted components. Thus, only DC components are extracted.  
         [0029]     According to the present invention, magnetic resonance signals can be received at higher sensitivity than heretofore.  
         [0030]     Other objects and features of the invention will appear in the course of the description thereof, which follows. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0031]      FIG. 1  is a conceptual diagram of digital quadrature detection performed in the present invention;  
         [0032]      FIG. 2  is a diagram illustrating lock-in detection;  
         [0033]      FIG. 3  is a diagram illustrating quadrature lock-in detection;  
         [0034]      FIG. 4  is a diagram illustrating digital lock-in detection;  
         [0035]      FIG. 5  is a block diagram of main portions of an apparatus according to the present invention;  
         [0036]      FIG. 6  is a flowchart illustrating one example of sequence of operations of the apparatus according to the present invention;  
         [0037]      FIG. 7  is a block diagram showing a first example of configuration of an NMR spectrometer; and  
         [0038]      FIG. 8  is a block diagram showing a second example of configuration of an NMR spectrometer. 
     
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0039]     Embodiments of the present invention are hereinafter described in detail with reference to the drawings.  
         [0040]      FIG. 5  is a block diagram showing main portions of an apparatus according to the present invention. An observed signal can be given by sin(ωt+α). The observed signal enters an ADC  60 , where the signal is converted into digital data. The digital data are passed as a sequence of digital numerical values into multipliers  61   a  and  61   b . Reference signals cos(ωt+β) and sin(ωt+β) are applied to the multipliers  61   a  and  61   b , respectively. In the multipliers  61   a  and  61   b , split parts of the observed signals are multiplied by the reference signals cos(ωt+β) and sin(ωt+β), respectively. The output signals from the multipliers  61   a  and  61   b  are passed through digital low-pass filters (LPFs)  62   a  and  62   b , respectively, so that high-frequency components are removed. As a result, the output signals from the filters  62   a  and  62   b  are DC signals sin(α−β) and −cos(α−β), respectively.  
         [0041]     The output from the multiplier  61   a  is given by sin(2ωt+α+β)+sin(α−β). On the other hand, the output from the multiplier  61   b  is given by cos(2ωt+α+β)−cos(α−β).  
       Embodiment 1  
       [0042]     The hardware configuration of an apparatus according to an embodiment of the present invention is similar to the configuration shown in  FIGS. 7 and 8  except that audio filters  18   a ,  18   b , A/D converters  31   a ,  31   b , signal processing portion  32 , and other components cope with sampling at high frequencies above approximately 2 MHz. Digital data accepted by the signal processing portion  32  are computed as shown in  FIG. 1 . The computation is performed serially every one NMR data set obtained by sampling. For this purpose, a high-speed board computer or digital signal processor (DSP) is required as the signal processing portion of relatively high computational speed. The signal processing portion is a device for processing data that the spectrometer receives. A serial detection NMR instrument can be implemented in this embodiment.  
         [0043]      FIG. 6  is a flowchart illustrating one example of sequence of operations of the apparatus according to the present invention.  
         [0044]     1) An NMR signal detected by the probe  6  enters the IRM (image reject mixer)  11  through the duplexer  5  and preamplifier  7 . In this mixer  11 , the NMR signal is mixed with a locally generated signal from the frequency synthesizer  20 . As a result, the output signal from the mixer has an intermediate frequency (IF). The signal from the mixer (IRM)  11  is passed through the filter  13  to pass only a given frequency band of 9 to 12 MHz. The IF signal produced from the digital direct synthesizer (DDS)  15  and passed through the DAC  16   b  is mixed with NMR signals in the mixer  17   b.    
         [0045]     Then, unwanted RF components are removed by the AF filter  18   b . The analog signal is converted into a digital signal by the ADC  31   b  (step S 1 ). The resulting data is temporarily held in a memory within the signal processing portion  32 . The processing is the same as the processing of digital quadrature detection already described in connection with  FIG. 4  up to this point.  
         [0046]     2) The obtained data is then stored in a first data buffer  40  ( FIG. 1 ) for one scan (step S 2 ). The data items are arrayed in the time-sequential order. The data set is copied to create two copy sets (hereinafter may be referred to as raw data). The first copy set is described below.  
         [0047]     The raw data are multiplied by cos Ωt in the time-sequential order and then passed through digital low-pass filter (D-LPF)  42   a.    
         [0048]     3) At this time, the cutoff frequency of the digital low-pass filter  42   a  is adjusted to be comparable to or lower than the digital resolution (e.g., 10 Hz) obtained by dividing the spectral width SW by the number of data points NP (step S 3 ). For example, where SW=10 kHz and NP=1000, the digital resolution is 10 Hz. The data sets are separately passed through a narrow-band digital low-pass filter (D-LPF)  49  (step S 4 ). A maximum intensity or time-averaged value of the obtained numerical values is stored in a data buffer  2 R. As a result, only signal intensities having the same frequency as Ω are stored in the data buffer  2 R.  
         [0049]     4) In the same way as in the above-described operations regarding the first data set, the raw data are multiplied by sin Ωt in the time-sequential order (step S 3 ). Then, the resulting data are passed through the digital low-pass filter  42   b  (step S 4 ). A maximum intensity or time-averaged value of the obtained numerical values is stored in the data buffer  2 I. As a result, the intensities of only signals which have a frequency equal to Ω and whose phase is different by 90° from the phase used in the above-described case are stored in the data buffer  2 I (step S 5 ).  
         [0050]     5) The above-described sequence of operations is then repeated as many times as a desired number of data points (NP times) with varied Ω (e.g., Ω−ΔΩ). During these operations, the amount of variation of Ω is NP×ΔΩ=SW. Control goes to step S 6 , where a decision is made as to whether the processing of step S 5  has completed (step S 6 ). If the result of the decision is affirmative (Yes), it follows that one process of FID (free induction decay) has been completed. If the decision is negative (No), the data point and Ω are incremented (step S 7 ). Control then returns to step S 3 . Thus, the intensities of signals which agree with frequencies from one end to the other end of the observation width SW can be recorded. Consequently, an NMR spectrum can be detected.  
         [0051]     In this way, according to the present invention, magnetic resonance signals can be received at higher sensitivity than heretofore.  
       Embodiment 2  
       [0052]     The hardware configuration of an apparatus according to the present embodiment of the invention is similar to the configuration shown in  FIGS. 7 and 8  except that audio filters  18   a ,  18   b , A/D converters  31   a ,  31   b , signal processing portion  32 , and other components cope with sampling at high frequencies above approximately 2 MHz. Digital data accepted by the signal processing portion  32  are computed as shown in  FIG. 1 . The computation is performed serially every one NMR data set obtained by sampling. For this purpose, a high-speed board computer or digital signal processor (DSP) is required as the signal processing portion of relatively high computational speed. The signal processing portion is a device for processing data that the spectrometer receives. A batch detection NMR instrument can be implemented in this embodiment. The sequence of operations of the instrument constructed in this way is described by referring to the flowchart of  FIG. 6 .  
         [0053]     1) An NMR signal detected by the probe  6  enters the IRM (image reject mixer)  11  through the duplexer  5  and preamplifier  7 . In this mixer  11 , the NMR signal is mixed with a locally generated signal from the frequency synthesizer  20 . As a result, the output signal from the mixer has an intermediate frequency (IF). The signal from the mixer  11  is passed through the bandpass filter  13  to pass only a given frequency band of 9 to 12 MHz. The IF signal produced from the digital direct synthesizer (DDS)  15  and passed through the DAC  16   b  is mixed with NMR signals in the mixer  17   b.    
         [0054]     Then, unwanted RF components are removed by the AF filter  18   b . The analog signal is converted into a digital signal by the ADC  31   b  (step S 1 ). The resulting data is temporarily held in a buffer within the signal processing portion  32 . The processing is the same as the processing of digital quadrature detection already described up to this point. Since the computational speed is low, oversampled data are once held. After the end of NMR experiments, the following calculations are performed.  
         [0055]     2) The obtained data is stored in a first data buffer  40  for one scan (step S 2 ). The data items are arrayed in the time-sequential order. The data set is copied to create two copy sets (hereinafter may be referred to as raw data).  
         [0056]     3) The first copy set is next described.  
         [0057]     The raw data are multiplied by cos Ωt in the time-sequential order (step S 3 ). Then, the resulting data are passed through the digital low-pass filter (D-LPF)  49  (step S 4 ). The cutoff frequency of the filter  49  is adjusted to be comparable to or lower than the digital resolution (e.g., 10 Hz) obtained by dividing the spectral width SW by the number of data points NP. For example, where SW=10 kHz and NP=1000, the digital resolution is 10 Hz.  
         [0058]     A maximum intensity or time-averaged value of the obtained numerical values is stored in the data buffer  2 R. As a result, only signal intensities having the same frequency as Ω are accumulated in the data buffer  2 R.  
         [0059]     4) In the same way as in the above-described session of operations 3), the raw data are multiplied by sin Ωt in the time-sequential order. Then, the resulting data are passed through the digital low-pass filter  42   b . A maximum intensity or time-averaged value of the obtained numerical values is stored in the data buffer  2 I (step S 5 ). As a result, the intensities of only signals which have a frequency equal to Ω and whose phase is different by 90° from the phase used in the above-described session 3) are accumulated in the data buffer  2 I.  
         [0060]     5) A decision is made as to whether the processing has been completed (step S 6 ). If the result of the decision is affirmative (Yes), it follows that one process of FID (free induction decay) has been completed. If the decision is negative (No), the above-described operations 3) and 4) are repeated as many times as a desired number of data points (NP times) with varied Ω (e.g., Ω−ΔΩ) (step S 7 ). During these operations, the amount of variation of Ω is NP×ΔΩ=SW. Thus, signal intensities which agree with frequencies from one end to the other end of the observation width SW can be recorded. Consequently, an NMR spectrum can be detected.  
         [0061]     In this way, according to the present invention, magnetic resonance signals can be received at higher sensitivity than heretofore.  
         [0062]     Having thus defined our invention with the detail and particularity required by the Patent Laws, what is desired protected by Letters Patent is set forth in the following claims.