Abstract:
A transmission suppression apparatus includes a first modulator that converts a symbol to a waveform. The apparatus further includes a first non-linear filter, configured to at least partially compensate for non-linear distortions of a transmission signal path. The apparatus further includes a first medium coupling device for coupling signals to a communication medium. The apparatus further includes a second medium coupling device for coupling signals from the communication medium. The apparatus further includes summing circuitry with a first input connected to an output of the second medium coupling device. The apparatus further includes cancellation circuitry, connected to a second input of the summing circuit, that converts the symbol to an analog waveform that is substantially 180 degrees out of phase with the analog waveform encoding the symbol on the first input to the summing circuit.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
       [0001]    This application claims priority to U.S. Application Ser. No. 61/114,845, filed on Nov. 14, 2008, which is incorporated herein by reference. 
     
    
     TECHNICAL FIELD 
       [0002]    This patent application relates generally to full-duplex digital communication, communication signal repeaters, and power line communications. 
       BACKGROUND 
       [0003]    In some cases it is desirable to have uninterrupted two-way communication between two devices. For example, telephone calls typically allow uninterrupted two-way communication to simulate a face-to-face communication. Two-way communication can be achieved by dedicating separate communication media to signals in each direction. In some circumstances using separate media can be expensive or impossible. Full-duplex communications allow two devices to both send and receive signals at the same time on a single communication medium. The problem that arises when the devices transmit and receive simultaneously on a communication medium is that the transmitted signal may interfere with the received signal and prevent accurate reception. One way to achieve full-duplex communications is to allocate different frequency bands to each direction of transmission. Confining the transmissions in each direction to non-overlapping frequency bands prevents the signals from interfering with one another. Band-pass filters may be used to cleanly receive each signal while alternate direction transmission continues. This approach may have some draw-backs in certain circumstances, such as reducing the usable bandwidth on the medium available for transmissions in each direction and thus limiting the rate of data transfer. 
         [0004]    Communications signal repeaters are devices used to relay signals between communication nodes that may share access to communication medium, but are still unable to communicate directly with each other because, for example, one node is out of range to reliably receive transmissions from another node. For example, repeaters are often employed in Power Line Communications (PLC) networks. Due to limited bandwidth (e.g., 2-80 MHz) and regulatory limits on radio frequency emissions, digital transmissions over power lines have limited range, typically 1-2 km. In order to propagate signals over longer distances on a power line, digital repeaters are mounted on pole tops at distances corresponding to the range limitations of the power line. Reaching customers located at the extreme end of a power line can require as many as 25 hops between repeaters. 
         [0005]    Transmissions requiring several hops can incur significant delay and consume a relatively large amount of available bandwidth because each retransmission of a signal occupies bandwidth on the communication medium. PLC devices typically share the medium via a Carrier Sense Multiple Access—Collision Avoidance (CSMA-CA) mechanism. This is essentially a listen-before-talk scheme. If the medium is busy, a station will wait until the medium is idle before sending any queued data. Transmissions on a CSMA-CA network are broken up into units of limited duration called frames. When a station has data to transmit and detects the medium is idle it will contend for the medium by commencing the transmission of a frame. If no collision occurs, the station will complete transmission of the frame. Upon completion of the frame, the station will relinquish the medium for at least a predetermined period of time to allow other stations to contend for the medium with transmissions of appropriate priority level. After the period of time expires, the station may commence transmission of another frame as needed. Repeaters must contend for the medium in order to retransmit a frame of data that they have received. Repeaters receive one or more complete frames of data and store that data until the repeater is able to successfully contend for the medium and commence retransmission of the data in a new frame or frames. This store and forward method causes an additional delay of at least the frame duration for each repeater hop in the path of a message. Each retransmission along the path also occupies bandwidth on the medium for the entire duration of the frame or frames. 
       SUMMARY 
       [0006]    In one aspect, in general, an apparatus includes a first modulator that converts a symbol to a waveform. The apparatus further includes a first non-linear filter, configured to at least partially compensate for non-linear distortions of a transmission signal path. The apparatus further includes a first medium coupling device for coupling signals to a communication medium. The apparatus further includes a second medium coupling device for coupling signals from the communication medium. The apparatus further includes summing circuitry with a first input connected to an output of the second medium coupling device. The apparatus further includes cancellation circuitry, connected to a second input of the summing circuit, that converts the symbol to an analog waveform that is substantially 180 degrees out of phase with the analog waveform encoding the symbol on the first input to the summing circuit. 
         [0007]    Aspects can include one or more of the following features. The first non-linear filter may be cascaded after the first modulator and the first medium coupling device may be cascaded after the first non-linear filter. The waveform may be a digitally encoded waveform. The apparatus may further include a first digital to analog converter cascaded after the first non-linear filter. The apparatus may further include a first analog amplifier connected to the output of the first digital to analog converter. The apparatus may further include a second analog amplifier with an automatic gain control circuit with the input connected to the output of the summing circuit. The apparatus may further include an analog to digital converter connected to the output of the second analog amplifier, a digital filter that converts the symbol to a digitally encoded waveform that is substantially 180 degrees out of phase with a residual signal encoding the symbol on the output of the analog to digital converter, and a digital summer that adds the output of the digital filter to the output of the analog to digital converter. The cancellation circuitry may include a second non-linear filter configured to at least partially pre-compensate for non-linear distortions of the cancellation circuitry. The first non-linear filter may be a memory polynomial filter. The communication medium may be a power line. The first non-linear filter may have a plurality of sets of coefficients, wherein each set of coefficients is associated with a different phase of the power cycle on the power line and each set of coefficients is adapted independently of the other sets of coefficients. The first modulator may be an orthogonal frequency division multiplexing modulator. The apparatus may further include a second analog amplifier connected to the output of the summing circuit, an analog to digital converter connected to the output of the second analog amplifier, and a signal path estimation block configured to estimate the non-linear distortion in the transmission signal path and the linear distortion in the transmission signal path based on the signal at the output of the analog to digital converter and the symbol. The signal path estimation block may be configured to estimate the linear distortion first and use the linear distortion estimate to estimate the non-linear distortion. The non-linear distortion estimate from the signal path estimation block may be used to configure the first non-linear filter and the estimation process is repeated with a subsequent symbol. The first medium coupling device and the second medium coupling device may share one or more common components. 
         [0008]    In another aspect, in general, a method includes filtering a transmission signal with a non-linear filter to at least partially pre-compensate for nonlinear distortion in a first signal path to generate a pre-compensated transmission signal. The method further includes coupling the pre-compensated transmission signal to a communication medium. The method further includes receiving, at a co-located receiver, an analog received signal from the communication medium that includes a component caused by the transmission signal. The method further includes filtering the transmission signal to generate an analog cancellation signal that is substantially 180 degrees out of phase with the component of the analog received signal that is caused by the transmission signal. The method further includes adding the analog cancellation signal to the analog received signal. 
         [0009]    Aspects can include one or more of the following features. The non-linear filter may be a digital filter. The method may further include converting the pre-compensated transmission signal to an analog transmission signal. The method may further include converting the received signal resulting from analog cancellation to a digital received signal, filtering the transmission signal to output a digital cancellation signal that is substantially 180 degrees out of phase with a residual component of the digital received signal that is caused by the transmission signal, and adding the digital cancellation signal to the digital received signal. Filtering the transmission signal to generate an analog cancellation signal may include filtering with a non-linear filter configured to substantially pre-compensate for non-linear distortions of a cancellation path. The non-linear filter may be a memory polynomial filter. The communication medium may be a power line. The transmission signal may be an orthogonal frequency division multiplexing signal. The communication medium may be a coaxial cable. The communication medium may be a twisted pair cable. An estimate of an impulse response of a signal path including the communication medium may be used to filter the transmission signal to generate an analog cancellation signal. 
         [0010]    In another aspect, in general, a method includes transmitting a plurality of orthogonal frequency division multiplexing symbols on a communication medium. The method further includes receiving the symbols from the communication medium at a co-located receiver. The method further includes applying a discrete Fourier transform to each of the received symbols to compute the frequency domain representation of the received symbols. The method further includes dividing the frequency domain representation of each of the received symbols by the frequency domain representation of the corresponding transmitted symbol. The method further includes averaging the quotients over all the symbols to estimate a transfer function of a first signal path. The method further includes dividing the frequency domain representation of each of the received symbols by the transfer function estimate and applying an inverse discrete Fourier transform to produce linear distortion compensated received symbols. The method further includes estimating the non-linear distortion in the first signal path based on the transmitted symbols and the linear distortion compensated received symbols. 
         [0011]    Aspects can include one or more of the following features. The method may further include configuring a non-linear filter to pre-compensate for non-linear distortions in the first signal path based on the estimate of the non-linear distortion in the first signal path, applying the non-linear filter to a plurality of orthogonal frequency division multiplexing symbols, and iterating the path estimation process, using the pre-compensated symbols to estimate the linear distortion of the signal path. The method may further include calculating the change in the linear and non-linear channel estimates from the last iteration and continuing to iterate until the change in the linear and non-linear channel estimates is below a threshold, at which point the linear and non-linear channel estimates are stored. The non-linear distortion on the first signal path may be modeled as a memory polynomial for estimation. The coefficient estimates for the memory polynomial may be calculated using a gradient descent algorithm. The gradient descent algorithm may use different adaptation step sizes for each harmonic branch of the memory polynomial. The gradient descent algorithm may use smaller step sizes for higher harmonic branches of the memory polynomial. The method may further include passing a plurality of orthogonal frequency division multiplexing symbols through a second signal path that includes a cancellation path to an analog summer in the co-located receiver and iterating the estimation process to estimate the linear and non-linear distortions in the second signal path. 
         [0012]    In another aspect, in general, a method includes transmitting a first multi-carrier signal on a communication medium and recovering a second multi-carrier signal from the communication medium, wherein the first multi-carrier signal and the second multi-carrier signal at least partially overlap in both frequency and time. Recovering the second multi-carrier signal includes adding a cancellation signal to a signal detected from the medium to suppress the first multi-carrier signal and recover the second multi-carrier signal. 
         [0013]    Aspects can include one or more of the following features. Recovering the second multi-carrier signal may include calculating the cancellation signal to be substantially 180 degrees out of phase with the component of the signal detected from the medium that is caused by the first multi-carrier signal. The communication medium may be a power line. The communication medium may be a coaxial cable. The communication medium may be a twisted pair cable. An estimate of an impulse response of a signal path including the communication medium may be used to filter the transmission signal to generate an analog cancellation signal. The first and second multi-carrier signals may be orthogonal frequency division multiplexed signals. The set of carrier frequencies modulated by data in the first multi-carrier signal and the set of carrier frequencies modulated by data in the second multi-carrier signal may intersect. The first multi-carrier signal and the second multi-carrier signal may be synchronized. Symbol boundaries of the first multi-carrier signal and the second multi-carrier signal may be aligned in time at the receiver. The first multi-carrier signal and the second multi-carrier signal may be synchronized so that symbol boundaries are aligned in time at the receiver. The multi-carrier signals may be broadband signals. The first multi-carrier signal may encode data from a frame that is still being received from the second multi-carrier signal. 
         [0014]    In another aspect, in general, an apparatus includes a modulator that converts a symbol to a digitally encoded waveform. The apparatus further includes a non-linear filter, configured to substantially pre-compensate for non-linear distortions of a transmission signal path, cascaded after the modulator. The apparatus further includes a digital to analog converter cascaded after the non-linear filter. The apparatus further includes an analog amplifier connected to the output of the digital to analog converter. The apparatus further includes a medium coupling device connected to the output of the analog amplifier. The apparatus further includes a receiver connected to the medium coupling device, receiving a detected signal appearing on a medium connected to the medium coupling device. The apparatus further includes a cancellation device that substantially cancels the representation of the symbol in the detected signal to determine a cancelled signal. The apparatus further includes an adaptation block that calculates new values for coefficients of the non-linear filter based in part on the cancelled signal. 
         [0015]    Aspects can include one or more of the following features. The cancellation device may include an analog summing circuit that is used to add a cancellation signal to the detected signal. The non-linear filter may be a memory polynomial filter. The medium coupling device may be connected to a power line and couple signals to and from the power line. The medium coupling device may be connected to a coaxial cable and couple signals to and from the coaxial cable. The medium coupling device may be connected to a twisted pair cable and couple signals to and from the twisted pair cable. The first non-linear filter may have a plurality of sets of coefficients, wherein each set of coefficients is associated with a different phase of the power cycle on the power line and each set of coefficients is adapted independently of the other sets of coefficients. The first modulator may be an orthogonal frequency division multiplexing modulator. 
         [0016]    In another aspect, in general, an apparatus includes a transmitter configured to modulate a first multi-carrier signal and couple the first multi-carrier signal to a communication medium. The apparatus further includes a receiver configured to couple signals from the communication medium and demodulate a second multi-carrier signal, wherein the first multi-carrier signal and the second multi-carrier signal at least partially overlap in both frequency and time. The apparatus further includes a processing device connected to both the transmitter and the receiver and configured to calculate a cancellation signal and add the cancellation signal to signals coupled from the communication medium by the receiver to suppress the first multi-carrier signal and recover the second multi-carrier signal. 
         [0017]    Aspects can include one or more of the following features. The cancellation signal may be substantially 180 degrees out of phase with the component of the signals coupled from the communication medium that is caused by the first multi-carrier signal. The communication medium may be a power line. The communication medium may be a coaxial cable. The communication medium may be a twisted pair cable. The processing device may calculate an estimate of an impulse response of a signal path including the communication medium that is used to filter the transmission signal to generate the cancellation signal. The cancellation signal may be analog and the receiver may include an analog summer circuit that is used to add the cancellation signal to signals coupled from the communication medium. The first and second multi-carrier signals may be orthogonal frequency division multiplexed signals. The set of carrier frequencies modulated by data in the first multi-carrier signal and the set of carrier frequencies modulated by data in the second multi-carrier signal may intersect. The first multi-carrier signal and the second multi-carrier signal may be synchronized. Symbol boundaries of the first multi-carrier signal and the second multi-carrier signal may be aligned in time at the receiver. The first multi-carrier signal and the second multi-carrier signal may be synchronized so that symbol boundaries are aligned in time at the receiver. The multi-carrier signals may be broadband signals. The transmitter may include a non-linear filter, configured to at least partially compensate for non-linear distortions of a transmission signal path. The non-linear filter may include coefficients that are calculated by the processing device based at least in part on signals coupled from the communication medium by the receiver. The processing device may calculate a second cancellation signal that is digital, the receiver may further include an analog to digital converter connected to the output of the analog summer circuit, and the second cancellation signal may be added to digital signals from the analog to digital converter. The second cancellation signal may be substantially 180 degrees out of phase with a residual component of the signals coupled from the communication medium that is caused by the first multi-carrier signal that remains after addition of the analog cancellation signal. The first multi-carrier signal may encode data from a frame that has been partially demodulated and is still being demodulated by the receiver from the second multi-carrier signal. 
         [0018]    In another aspect, in general, an apparatus includes a means for transmitting a first multi-carrier signal on a communication medium. The apparatus further includes a means for recovering a second multi-carrier signal from the communication medium, wherein the first multi-carrier signal and the second multi-carrier signal at least partially overlap in both frequency and time. Recovering the second multi-carrier signal includes adding a cancellation signal to a signal detected from the medium to suppress the first multi-carrier signal and recover the second multi-carrier signal. 
         [0019]    Among the many advantages of the invention (some of which may be achieved only in some of its various aspects and implementations) are the following. 
         [0020]    Transmission signals from a transmitter that are detected by a collocated receiver can be suppressed to enable simultaneous transmission and reception of signals on a communication medium while reusing bandwidth for both transmission and reception. For example, the described methods and apparatus may be applied in a full-duplex orthogonal frequency division multiplexed (OFDM) communications system with upstream and downstream signals simultaneously occupying some or all of the same frequency spectrum. In another example, the described methods and apparatus may be applied in a communications signal repeater to reduce delay through a network without reducing throughput by allowing the repeater to begin the forwarding transmission of a frame of data before reception of the entire frame is completed, while reusing some or all of the same frequency spectrum on the communication medium. Furthermore, some of the methods and apparatus described in this application provide for effective local transmit cancellation in the presence of transmit amplifiers and transmission channels that have both linear and non-linear distortion components. 
         [0021]    The transmit signal is pre-compensated for non-linear distortion such that the signal, when it reaches the communication medium, is cleaner in the sense that the non-linear distortions of the transmit amplifier are mitigated. By adding a cancellation waveform to the received waveform in the analog domain, the signal-to-noise ratio at the digital receiver is improved and the requisite dynamic range or required bits of the analog-to-digital converter are reduced. Estimating the linear distortion effects independently from the non-linear distortion effects, yields a highly computationally efficient cancellation model. The system allows for robust full duplex communication on a network with large distortions, attenuations and reflections that result in a reflected version of the transmit signal with power that is large relative to signal received from a remote transmitter. 
         [0022]    Some of the foregoing method(s) may be implemented as a computer program product comprised of instructions that are stored on one or more machine-readable media, and that are executable on one or more processing devices. The foregoing method(s) may be implemented as an apparatus or system that includes one or more processing devices and memory to store executable instructions to implement the method. 
         [0023]    The details of one or more examples are set forth in the accompanying drawings and the description below. Further features, aspects, and advantages of the invention will become apparent from the description, the drawings, and the claims. 
     
    
     
       DESCRIPTION OF THE DRAWINGS 
         [0024]      FIG. 1  is a block diagram of a digital communication transceiver employing transmission suppression. 
           [0025]      FIG. 2  is a block diagram showing signal paths and their associated distortions. 
           [0026]      FIG. 3  is a block diagram of a data modulator used to generate an analog transmit-cancellation signal. 
           [0027]      FIG. 4  is a flowchart of a process for computing estimates of the non-linear and linear channel characteristics of a signal path. 
           [0028]      FIG. 5  is block diagram a TX distortion measurement block. 
           [0029]      FIG. 6  is schematic of a memory-polynomial based pre-distorter used for non-linearity compensation. 
           [0030]      FIG. 7  is a detailed block diagram of a digital communication transceiver employing transmission suppression. 
           [0031]      FIG. 8  is a schematic diagram of a transmission line with repeaters. 
       
    
    
       [0032]    Like reference numerals in different figures indicate like elements. 
       DETAILED DESCRIPTION 
       [0033]    There are a great many possible implementations of the invention, too many to describe herein. Some possible implementations that are presently preferred are described below. It cannot be emphasized too strongly, however, that these are descriptions of implementations of the invention, and not descriptions of the invention, which is not limited to the detailed implementations described in this section but is described in broader terms in the claims. 
         [0034]    Cancellation techniques are used to achieve full-duplex communication or low-delay forwarding on a communication medium while allowing the transmitted and received signals to overlap in both time and frequency. When the transmitted signal is coupled to the medium, it is detected by a collocated receiver along with a desired signal from a remote device. A cancellation signal is generated based upon information about the transmitted signal and the signal paths. The cancellation signal is added to the detected signal to substantially suppress or eliminate the components associated with the transmitted signal and facilitate reception of the desired signal. The cancellation signal may be generated by applying adaptive linear and/or non-linear filters to a representation of the transmitted signal. 
         [0035]    Non-linearities in the transmit signal path may be pre-compensated using a adaptive non-linear filter in the transmit path, thus simplifying the adaptation of the filters in the cancellation signal path. The non-linear pre-compensation filter may be adapted based on measurements of the detected signal, possibly after processing to remove linear distortions from the signal. The use of a non-linear pre-compensation filtering in the transmit path has the additional benefit of providing a cleaner transmitted signal on the medium, thus facilitating remote reception and regulatory compliance. 
         [0036]    Some or all of the cancellation may be performed on the analog detected signal by using an analog summing circuit to apply an analog cancellation signal to the detected signal prior to analog to digital conversion. Applying a cancellation signal in the analog domain may allow the use of a digital to analog converter with a smaller dynamic range which is generally cheaper. This advantage is most pronounced when the power of the component of the detected signal corresponding to the transmitted signal is large relative to the power of the desired signal component. A digital cancellation signal may be applied to the received signal after analog to digital conversion to further suppress any residual components in the signal relating to the transmitted signal. 
         [0037]    Some implementations of the physical (PHY) layer use OFDM modulation. In OFDM modulation, data are transmitted in the form of OFDM “symbols.” Each symbol has a predetermined time duration or symbol time T s . Each symbol is generated from a superposition of N sinusoidal carrier waveforms that are orthogonal to each other and form the OFDM carriers. Each carrier has a center (or “peak”) frequency f i  and a phase Φ i  measured from the beginning of the symbol. For each of these mutually orthogonal carriers, a whole number of periods of the sinusoidal waveform is contained within the symbol time T s . The symbol time T s  does not include added time between symbols for features of a transmission protocol such as a guard band or cyclic prefix. Equivalently, each carrier frequency is an integral multiple of a frequency interval Δf=1/T s . The phases Φ i  and amplitudes A i  of the carrier waveforms can be independently selected (according to an appropriate modulation scheme) without affecting the orthogonality of the resulting modulated waveforms. The carriers occupy a frequency range between frequencies f 1  and f N  referred to as the OFDM bandwidth. 
         [0038]      FIG. 1  depicts an exemplary transmit cancellation system where OFDM symbol data  10 , generated by a micro controller or other such data source, is passed to the TX Data Modulator  11 . The TX Data Modulator  11  digitally transforms the symbol data into a corresponding digitally represented spectrally encoded OFDM symbol. The TX Data Modulator  11  then converts this frequency domain symbol into a digitally encoded time domain symbol, and adds an appropriately sized guard interval to the time domain waveform. The digitally encoded time domain waveform data is then passed to the non-linear pre-compensation block  18  where the inverse of the nonlinearities of the system&#39;s transmit and receive signal propagation path H 1   36  ( FIG. 2 ), which has been measured by the TX Distortion Measurement block  13 , are applied to the transmit waveform. The digitally represented pre-compensated time domain symbol waveform data is then passed to a digital-to-analog converter (DAC)  16  where it is translated into a time domain voltage waveform. This voltage waveform is then amplified in the transmit amplifier TX AMP  20  to an appropriate power level and coupled by the coupler  23  to the communication medium  25  where the waveform will be observable by all receivers within range, including the local receiver which is co-located with the transmitter blocks. The transmitted signal that enters the co-located receiver may cause interference with the reception of a signal from a distant transmitter. The cancellation system is able to suppress any such interference. 
         [0039]    The waveform used to cancel the transmitted waveform as it appears at the receive summer  22  is also computed from the same OFDM symbol data  10  used by the TX Data Modulator  11 . Much like the TX Data Modulator  11  the Cancellation (CX) Data Modulator  12  digitally transforms the symbol data into a corresponding digitally represented spectrally encoded OFDM symbol. This spectrally encoded symbol is then adjusted for the system&#39;s linear distortions by multiplying it&#39;s spectrally encoded representation by the spectral representation of the composite linear channel distortion computed by the TX Distortion Measurement block  13 . This composite linear channel distortion is equal to the linear spectral distortion experienced by path H 1   36  divided by the linear spectral distortion experienced by path H 2   37 . These distortions are measured by the TX Distortion Measurement block  13  and stored (e.g., in a memory within the block  13 ). After the spectrally encoded symbol has been compensated for the composite linear distortions of the system, it is transformed into the time domain, where the appropriate length guard interval is added to its time domain digital representation. Let h 1  and h 2  refer to the impulse response of the two paths H 1  and H 2  respectively. Let x refer to the spectrally encoded OFDM symbol in the time domain at the output of the TX Data Modulator  11 . Then the spectrally-encoded cancellation OFDM symbol in the time-domain, x c , is given by 
         [0000]        x   c   =F   N   −1 ( F   N ( x )* F   N ( h 1)/ F   N ( h 1)) 
         [0040]    In the above equation, N represents the Fast Fourier Transform (FFT) size used in the OFDM system, and F N  represents the N-point FFT operation. The inverse FFT (F N   −1 ) operation above represents the N-point IFFT as defined by the system and could involve the conjugate symmetric extension of the argument. 
         [0041]    Note that h 1  and h 2  or F N  (h 1 ) and F N  (h 2 ) may be computed and stored in the TX distortion measurement block  13 . 
         [0042]    The time domain cancellation signal (x c ) in the above equation is then extended with a corresponding prefix for the guard interval. Thus, in essence, the CX data modulator filters the output of the TX data modulator  11  and performs the same operation as the TX data modulator on this filtered output. The described operation of the CX data modulator is depicted in  FIG. 3 . The filter  46  in  FIG. 3  is described in the frequency domain in the above equation. The system may reuse hardware by sharing building blocks, such as FFT engines, in the various digital signal paths for a more cost effective and efficient implementation. Though  FIG. 1  shows one embodiment where the CX data modulator  12  receives its input from the output of the TX data modulator  11 , the CX data modulator could alternatively receive the input symbol data  10  directly in order to compute the cancellation signal. 
         [0043]    The time domain signal at the output of the CX data modulator  12  is then pre-compensated for the nonlinearities experienced by path H 2   37 . These nonlinearities are measured by, and stored in, the TX Distortion Measurement  13  block and applied to the digitally represented time domain symbol by the CX Non-linear Pre-compensation  26  block. After the digitally represented time domain symbol has been fully compensated it is passed to the CX DAC  17  where it is translated into a corresponding analog voltage waveform. It should be noted that the time domain symbol transmitted from the CX DAC  17  is transmitted synchronously with the corresponding symbol which is transmitted from the TX DAC  16 . The analog voltage waveform coming from the CX DAC  17  is then amplified by the CX AMP  21  which drives the analog summer  22 . After the summing process, the signal leaving the analog summer  22  and driving the RX AMP  24  will contain all the signals found on the medium, with the exception of the all or part of the signals transmitted by the local transmitter (e.g., from any of the blocks  11 ,  18 ,  16 , or  20 ) which have been cancelled out in the summation process. Any residual signals from the output of the TX AMP  20  that remain at the output of the RX AMP  24  will be removed in the Digital TX signal cancellation block  14 . 
         [0044]    In OFDM systems, the complexity of the transmission suppression system may be reduced by synchronizing the transmitted OFDM signal with the received OFDM signal to exploit the guard interval, or cyclic prefix. Synchronization allows the cancellation filtering to be performed on a symbol by symbol basis using cyclic convolution, instead of performing a more complex linear convolution on the sequence of OFDM symbols. When symbol by symbol cyclic convolution is used errors are created in the cancellation signal at the symbol boundaries due to the inaccuracy of the approximation of the transmitted signal as a periodic signal. These errors have a duration determined by the length of the impulse response of the channel estimate. In this case it is desirable to have a guard interval that is at least as long as the delay spread of the channel H 1 . When this condition on the guard interval is met and the symbols of the two OFDM signals are aligned at the receiver, the error in the cancellation signal occurs during the guard interval of the desired received signal, which is discarded by the receiver anyway. Thus, the cancellation approach described above exploits the guard interval in multicarrier systems to avoid cancellation at symbol boundaries, thereby greatly simplifying the cancellation process. 
       Measurement Process—Analog Cancellation Loop 
       [0045]    In order for the aforementioned analog transmit power cancellation process to provide accurate cancellation, accurate linear and non-linear measurements of the transmit path H 1   36  and the cancellation path H 2   37  should be made and stored. This can be done, for example, in the following manner. 
         [0046]    As the transmitted signal propagates from the TX DAC  16  to the RX ADC  19  along path H 1   36 , it experiences numerous linear and non-linear distortions.  FIG. 2  shows a partial block diagram that depicts some representative distortions experienced by the signal that is transmitted from the TX DAC  16  as it propagates along propagation path H 1   36  to the RX ADC  19 , and some representative distortions  34  and  35  experienced by the cancellation signal transmitted from the CX DAC  17  as it follows path H 2   37  to the RX ADC  19 . Distortions caused by the TX AMP  20 , CX AMP  21 , and RX AMP  24  are represented by replacing those blocks with TX Dist block  32 , Cancel Dist block  34  and RX Dist block  35 , respectively. Distortions caused by the coupler  23  and communication medium  25  are represented as a single Coupler &amp; Medium Dist block  33 . These distortions are varied in their nature and may be caused by these or other parts of the system in different proportions. In the illustrated example, the major source of non-linear distortion for path H 1   36  is generated by the transmit amplifier TX AMP  20  and the major source of non-linear distortion for path H 2   37  is generated by the cancellation amplifier CX AMP  21 . Additionally, in this example the major source of linear distortion for path H 1   36  is most often due to the effects of the communication medium  25  as it is coupled via the coupler  23  to the signal path. In other words, a reflected version of the transmitted signal travels through the channel before entering the co-located receive port leading to a linear distortion that is caused by the channel. 
         [0047]    Accurate measurements of the linear and non-linear distortions in paths H 1   36  and H 2   37  can be attained by using the method described in the flowchart of  FIG. 4 . The training process for measuring the linear and non-linear distortion of path H 1   36  will be used as an example. Similar techniques can also be employed to measure the distortion of path H 2   37 . Referring to  FIG. 7 , during the training phase, switch  28  remains in the up position connecting the Rx ADC  19  to the TX distortion measurement block  13 . When the linear and non-linear distortions on path H 1  are being measured, switches  70  and  72  are closed and switch  71  is open. When measuring distortions on path H 2 , switches  71  and  72  are closed and switch  70  is open. To start the measurement process the channel under measurement, in this case H 1   36 , is assumed to be completely linear  51 , hence no non-linear pre-compensation is applied to the initial signal to be transmitted. First, a multi-symbol training waveform  52  is generated and transmitted  53  through path H 1   36 . Each symbol received at the RX ADC  19  is moved into the frequency domain and divided by the corresponding spectrum of the same symbol before it was transmitted (the undistorted spectrum of the original symbol). The quotient of this per-symbol division operation is then averaged over a sufficiently large number of symbols. The averaging process spreads the power of the noise and other uncorrelated signals and increased the accuracy of the linear channel distortion estimate  54 . The estimate of the linear distortion  54  is then saved for later use. 
         [0048]    Let N s  OFDM symbols be used to estimate H 1 . Let x i  represent the time-domain OFDM symbol ‘i’ at the output of the TX data modulator  11 , and let y i  represent the corresponding received symbol at the input of the RX ADC  19 . Then, the computation of the channel&#39;s linear spectral transformation (distortion) described above can be written as follows: 
         [0000]        H 1=(1 /N   s )*Σ i   F   N ( y   i )/ F   N ( x   i ), i=1, 2, . . . , N s    
         [0049]    After the linear channel distortion  54  has been computed, the effects of the linear channel are then removed from one or more of the received symbols  55  by dividing the spectral description of the received symbol by the estimated channel (H 1 ). 
         [0000]        z   i   =F   N   −1 ( F   N ( y   i )/ H 1) 
         [0050]    The linearly compensated RX symbol z i  is then used to compute the inverse non-linearity  56  of the signal path being measured, which is, in this case, H 1   36 . Note that for signal path H 1 , x i  is the input and z i  represents the non-linear output (because the effect of the linear component of the channel has been removed in the computation of z i ). Thus, 
         [0000]        x   i   =G ( z   i ), 
         [0000]    where G represents the inverse non-linearity function. The two quantities x i  and z i  are used to adaptively estimate the inverse non-linearity function G. The procedure to estimate G is presented later. The operation of the TX distortion measurement block  13  as described in  FIG. 4  is shown in more detail in  FIG. 5 . In this embodiment, an adaptive algorithm is used to estimate G in block  56 . 
         [0051]    A new multi-symbol calibration waveform is then generated as before. This waveform is then moved into the digital time domain where it is pre-compensated for the systems nonlinearities  58  using the inverse non-linearity function, G, estimated by the non-linear distortion measurement block  18 . Suppose the calibration waveform consisted of a sequence of time-domain OFDM symbols {α i }, the pre-compensated transmit waveform is given by {G (α i )}. 
         [0052]    The pre-compensated waveform is then transmitted  53  and received as before. The received linearized (pre-compensated) waveform is then used to compute a more accurate estimate of the linear channel characteristics  54 . The new, more exact, linear channel estimate is then removed from the signal, and, as before, the resultant signal is used to estimate the non-linear channel characteristics  55  and  56 , which can again be used to transmit another, more accurately pre-compensated  58  channel calibration waveform. This process is repeated until the accuracy of the linear and non-linear channel estimates are adequate for the application  57 , at which point the linear and non-linear channel estimates are stored for later use  59  and the calibration process is stopped  60 . 
       Measurement and Cancellation Process—Digital Loop 
       [0053]    Due to imperfect measurements and imperfect device characteristics, the transmit power cancellation achieved at the analog summer  22  may be less than required for optimum performance. In order to further improve the removal of the transmitter&#39;s power from the received signal, digital cancellation loops can be implemented. These loops can include linear cancellation loops and/or non-linear cancellation loops. 
         [0054]      FIG. 2  shows a representative digital linear cancellation loop. The remaining channel distortion H 3   40  is computed by taking the spectrum of the OFDM symbol received at the RX ADC  19  and dividing it by the original symbol spectral data Symbol Data  10  which was used by the transmitter when generating the symbol now being received. This value is then averaged over a number of symbols to improve its accuracy and to spread the power of noise and interfering signals. The computed remaining linear cancellation distortion H 3   42  is then multiplied by the negative of the original symbol&#39;s spectral data Symbol Data  10  yielding the inverse of the transmitted symbols remaining power. This cancellation spectral power is then added  43  to the spectrum of the received symbol, thereby further reducing the transmitted symbols power found in the received symbol data RXD  44 . The process is very similar to the one used to estimate H 1  and H 2  as described above. Referring to  FIG. 7 , during the training phase for estimating H 3 , switch  28  is in the up position, switch  70  and  71  are closed, and switch  72  is open. Thus, the distortion measured by the TX distortion block  13  is the residual linear distortion (H 3 ) after non-linearity pre-compensated transmission and analog transmit signal cancellation. 
         [0055]    For additional system performance digital non-linear cancellation loops can also be implemented. This digital non-linear cancellation will work in conjunction with the digital linear cancellation loop much like the non-linear estimation and cancellation process described in  FIG. 4  and explained in the analog cancellation section. 
         [0056]    Note that switches  70 , 71 ,  72 , and  25  that are shown in  FIG. 7  are only present to simplify exposition and to identify the path of signal-flows during different training and calibration modes of operation. Any implementation need not have any or all of these switches. These switches can be replaced with short-circuits and necessary paths can be turned on and off digitally. 
       Estimating the Inverse Non-Linearity Function 
       [0057]    As mentioned earlier, training symbols are used to estimate the linear and non-linear components of the signal transmission path. Let x i  be a transmitted OFDM symbol, and z i  be the corresponding non-linear component at the output of the transmission path. In other words, z i  is the received symbol from which the effects of the linear channel has been removed in block  55 . It has been said before that the relationship between x i  and z i  can be expressed as x i =G(z i ), where G represents the non-linear component of the transmission path. 
         [0058]    The inverse non-linearity is modeled using memory polynomials (also known as non-linear tapped delay lines). Thus, the relationship between x i  and z i  explicitly be expressed as 
         [0000]        x   i ( n )=Σ k Σ q   w   kq   z   i ( n−q )| z   i ( n−q )| k−1   , q= 0, 2 , . . . , Q− 1, and k⊂{1, 2, 3, . . . }. 
         [0059]    In the above equation, k is the set of harmonics that we are trying to suppress, and Q−1 is the memory of the system.  FIG. 6  illustrates the memory polynomial model of the non-linearity with Q=2 and k={1, 2, 3, 5}. During the training process, x i  and z i  are used to compute the weights w kq    61 . The weights are obtained using a gradient descent algorithm like the LMS (least-mean-squares) algorithm. In one embodiment of the algorithm that uses LMS, the weights are obtained in an iterative manner as follows: 
         [0000]        w   kq ( n+ 1)= w   kq ( n )+μ k   z   i ( n−q )| z   i ( n−q )| k−1 ( x   i ( n )−Σ k Σ k   w   kq ( n ) kq ( n ) z   i ( n−q )| z   i ( n−q )| k−1 ), 
         [0000]    where μ k  represents the step size that is used to adapt the coefficients corresponding to the kth harmonic viz., w kq , q=0, 2, . . . , Q−1. 
         [0060]    In one embodiment of the algorithm, every harmonic branch uses a different step size for faster convergence. Once the weights w kq  are determined, the inverse non-linearity function is fully defined, and it can be used for non-linear pre-compensation on the transmit path. 
         [0061]    A transceiver employing the transmission suppression method described above may be used by a communication signal repeater to reduce forwarding delay and enhance network throughput. An exemplary repeater application in a PLC network is depicted in  FIG. 8 . The PLC network  800  includes a head end  810  and several stations (e.g.  811 ,  812 ,  813 ,  814 ,  815 ,  816 ,  817 , and  818 ) operating repeaters and positioned on poles spaced along the power line  820  such that only adjacent stations are within reception range of each other. The repeaters include transceivers with the transmission suppression capabilities described above. In the example scenario the head end  810  has data to transmit to station  815 . Head end  810  first partitions the data into one or more CSMA-CA frames and sets the destination field for the frame or frames to the address for station  815 . Head end  810  may also set the value of a control field in the frame header to indicate that immediate forwarding is enabled. 
         [0062]    When head end  810  detects that the medium is idle, it transmits the first frame on the power line  820  using a PHY layer protocol such as, for example, OFDM. The repeater at station  811  begins reception of the frame and checks the destination address. Because station  811  is not the destination and the immediate forwarding is enabled, the repeater begins copying the incoming frame and commences retransmission of the frame before reception of the frame is complete. As it retransmits the frame, station  811  may clear the immediate forwarding control field to indicate that immediate forwarding for the next hop is disabled. Head end  810  is still transmitting the first frame and ignores the retransmission. Station  812  then begins reception of the retransmitted frame and engages its own repeater. Because the immediate forwarding is disabled, station  812  stores the frame until reception is complete and then commences retransmission of the frame. As it retransmits the frame, station  812  may set the immediate forwarding control field to indicate that immediate forwarding for the next hop is enabled. This process of reception and retransmission continues at each repeater  813  and  814  down the power line  820  until station  815  receives the retransmission of the frame from the repeater at station  814 . Station  815  checks the destination address for the frame and determines that it is the final destination of the frame. Thus station  815  does not retransmit the frame, completes reception and decode of the frame so the payload may be passed up for higher network layer stack processing at station  815 . 
         [0063]    After head end  810  completes transmission of the first frame it will wait a length of time sufficient allow retransmission of the first frame by a non-adjacent station, in this case station  812 , or until an acknowledgment for the first frame is received. Head end  810  will then attempt to contend for the medium  820  in order to transmit the next remaining frame if any. The entire process will be repeated until all frames have been sent from head end  810  and received by destination station  815 . 
         [0064]    The amount of time the head end  810  must wait after completion of its transmission of the first frame to start transmission of the next frame is reduced compared to a system that stores the entire frame before commencing retransmission from the repeater at station  811 , because a substantial portion of the frame may be retransmitted prior to completion of the first transmission. Thus head end  810  is able to transmit a sequence of frames faster and a higher network throughput is achieved by reusing the bandwidth on the medium  820  for simultaneous forwarding. The system may also achieve a higher data rate than a comparable system using non-overlapping frequency bands for the transmission and retransmission of a forwarded frame, because more of the usable bandwidth on the medium  820  may be used for each transmission. 
         [0065]    Repeaters employing transmission suppression may reuse bandwidth used by an incoming transmission for concurrent retransmission as long as the destination node (e.g. the next repeater in the repeater chain or the ultimate destination node) is sufficiently remote from the source node (e.g. the previous repeater in the chain or the ultimate source node). If the source node is sufficiently remote from the destination node then interference from the incoming transmission will be small enough to allow reliable reception of the retransmission at the destination node. In this manner forwarding delay is reduced relative to a store and forward repeater scheme while data rates and network throughput may be kept high by efficiently reusing some or all available bandwidth on the medium. 
         [0066]    Any processes described herein and their various modifications (hereinafter “the processes”), are not limited to the hardware and software described above. All or part of the processes can be implemented, at least in part, via a computer program product, e.g., a computer program tangibly embodied in an information carrier, such as one or more machine-readable media or a propagated signal, for execution by, or to control the operation of, one or more data processing apparatus, e.g., a programmable processor, a computer, multiple computers, and/or programmable logic components. 
         [0067]    A computer program can be written in any form of programming language, including compiled or interpreted languages, and it can be deployed in any form, including as a stand-alone program or as a module, component, subroutine, or other unit suitable for use in a computing environment. 
         [0068]    Actions associated with implementing all or part of the processes can be performed by one or more programmable processors executing one or more computer programs to perform the functions of the calibration process. All or part of the processes can be implemented as, special purpose logic circuitry, e.g., an FPGA (field programmable gate array) and/or an ASIC (application-specific integrated circuit). 
         [0069]    Processors suitable for the execution of a computer program include, by way of example, both general and special purpose microprocessors, and any one or more processors of any kind of digital computer. Generally, a processor will receive instructions and data from a read-only memory or a random access memory or both. Components of a computer include a processor for executing instructions and one or more memory devices for storing instructions and data. 
         [0070]    Components of different embodiments described herein may be combined to form other embodiments not specifically set forth above. Other embodiments not specifically described herein are also within the scope of the following claims.