Abstract:
A voltage, replica of the difference between two dissimilar base-emitter voltages in the form of an intrinsic input offset voltage of a differential input pair of transistors of a noninverting, buffer-configured operational amplifier, is summed with a pre-established fraction of a base-emitter voltage, to produce a voltage reference without thermal drift of a level that can be as low as few 10 mV. The intrinsic input offset voltage is controlled by a local feedback loop acting on the bias current that is forced through the input pair of transistors that may be realized with a certain area ratio. The relatively simple circuit is useful in battery operated, low supply voltage, systems.

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This application is a continuation of application Ser. No. 08/358,159 filed Dec. 16, 1994 and now abandoned. 
    
    
     This application claims priority from EP 93830512.5, filed Dec. 17, 1993, which is hereby incorporated by reference. 
     BACKGROUND AND SUMMARY OF THE INVENTION 
     The present invention relates to a method and a circuit for generating a reference voltage without thermal drift and of relatively low value, i.e. markedly lower than the voltage of a base-emitter junction (Vbe). 
     In many systems and particularly in monolithically integrated systems, it is necessary to implement voltage references, that is circuits capable of generating a stable reference voltage, free of thermal drift. Commonly this is achieved by employing a so-called band-gap circuit. A band-gap circuit produces a voltage corresponding to the sum of one or several base-emitter voltages (Vbe), as of common bipolar junction transistors, and of a voltage proportional to the difference between two different base-emitter voltages, suitably amplified by a certain amplification factor K, so as to make the amplified difference voltage comparable with the voltage of one or several base-emitter junctions, in order to produce a desired reference voltage given by: 
     
       
           V ref= Vbe+KδVbe , where  K&gt; 1. 
       
     
     The ΔVbe term that is employed for compensating the thermal drift of a certain sign of the particular Vbe or sum of Vbe used, may suitably assume a thermal coefficient of opposite sign of the thermal coefficient of the Vbe term used. Therefore, the resulting reference voltage Vref that is produced may be stable in terms of temperature variations. 
     Commonly band-gap circuits produce a temperature compensated voltage Vref greater or equal to about 1.2V. On the other hand, in systems designed for operating with relatively low supply voltages, for example in battery powered portable instruments and apparatuses, the supply voltage may be relatively low, for example in the vicinity of 1.0V. This makes a correct operation of a normal band-gap circuit impossible. 
     Recently, a band-gap reference voltage generating circuit has been proposed which is capable of providing a regulated voltage of relatively low level, in the vicinity of 200 mV, which may be adjusted upward to higher levels. This makes the voltage reference circuit suitable also in battery powered systems with a supply voltage of just 1V. The circuit is described in the article entitled: “A Curvature-Corrected Low-Voltage Bandgap Reference”, by Gunawan, Meijer, Fonderie, and Huijsing, 28 IEEE JOURNAL OF SOLID STATE CIRCUITS 667-670 (1993), the content of which is herein incorporated by express reference. 
     Such a known circuit adopts a compensating system of the nonlinearity of the temperature characteristics of a base-emitter junction (Vbe). Basically, the circuit employs a first circuit block for generating a current proportional to the absolute temperature (PTAT) and a second circuit block capable of generating a current proportional to a Vbe, plus a correction current for compensating the nonlinearity of the temperature coefficient of the Vbe. Thereafter, the sum of the two currents is converted to a voltage signal which is amplified by an output buffer. The circuit is relatively complex and generates a stabilized reference voltage of about 200 mV, with a supply voltage that may be as low as about 1V. 
     There remains a need or utility for a circuit capable of generating a reference voltage of a relatively low value (on the order of a few tens of mV) without thermal drift, which is relatively simple to realize. 
     This objective is fully met by the method and the circuit object of the present invention. 
     Basically, the method of the invention rests on the generation of a stabilized voltage in the form of a sum of a voltage equivalent to the difference between two different base-emitter voltages, which is advantageously represented by a suitably controlled intrinsic offset voltage of a pair of transistors that constitute an input differential stage of a buffer-configured, operational amplifier, and a preestablished fraction of a base-emitter junction voltage. A subdivision of a Vbe voltage is implemented by mirroring, in a certain ratio, a current proportional to a Vbe voltage and by converting the divided-down mirrored current into a divided-down Vbe voltage on a resistance. The voltage difference between two different base-emitter junction voltages to be summed with the divided-down portion of a Vbe voltage, in order to compensate in terms of temperature the resulting voltage sum, is obtained in the form of an intrinsic offset voltage, controlled through a local feedback loop, of a differential pair of transistors that form an input stage of an operational amplifier that practically works as an output buffer of the stabilized voltage produced by the circuit. 
     The stabilized voltage sum that can be generated by the circuit may be of several tens of milliVolts and may be freely scaled-down by the use of a resistive voltage divider. 
     The circuit may be powered with a voltage of about 1V without jeopardizing its operation. Therefore the circuit is particularly useful in low voltage, battery powered systems. 
    
    
     BRIEF DESCRIPTION OF THE DRAWING 
     The disclosed inventions will be described with reference to the accompanying drawings, which show important sample embodiments of the invention and which are incorporated in the specification hereof by reference, wherein: 
     FIG. 1 is a diagram of a circuit for generating a reference voltage, according to the present invention; 
     FIG. 2 is a partial, simplified circuit diagram of the circuit of FIG. 1, which emphasizes some essential aspects of the circuit of the invention; 
     FIG. 3 shows a voltage-temperature characteristic of a circuit made according to the present invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     The numerous innovative teachings of the present application will be described with particular reference to the presently preferred embodiment (by way of example, and not of limitation), in which: 
     With reference to FIG. 1, the portion on the circuit of the left-hand side of the node V 1  will be discussed first. 
     The bipolar junction transistor (BJT) Q 2  generates a current I given by the ratio between its base-emitter voltage Vbe and the resistance R 1 : I=Vbe Q2 /R 1 . 
     A suitable start-up circuit may comprise, as shown, a current generator I1, which in practice may be constituted by a transistor Q 0  of an appropriate size. As a matter of fact, a so-called start-up circuit is necessary in order to ensure that, at the turn-on instant, the local loop reaches a self-sustaining condition. 
     To a first approximation, the base current of the BJT Q 2  under equilibrium conditions, will be given by: Ic Q2 =I1. This condition will then be maintained stable by the local feedback loop. However, at the turn-on instant, Q 2  is still off and will turn-on only when Ic Q4 =Vbe Q2 /R 1 . Therefore, the collector voltage of Q 2  will tend to drop until Q 1  (which triggers the feedback) turns on thus supplying a current to Q 3 , which current, mirrored by Q 4 , drives the base of Q 2 . This driving current of the BJT Q 2  will continue to increase until the following relationships hold: 
     
       
           Ic   Q1   =Ic   Q3   =Ic   Q4   =Vbe   Q2   /R 1. 
       
     
     The transistor Q 5 , having the same area as Q 1 , will also conduct a current given by: Ic Q5 =Vbe Q2 /R 1 , which will be forced on R 2 , thus producing the voltage signal V 1 . The resistances RE Q1  and RE Q5 , which should be equal, serve for degenerating the respective current generators Q 1  and Q 5 . Therefore, a fraction of the Vbe Q2  voltage, given by 
     
       
           V   1 = Vbe   Q2   *R 2/ R 1 
       
     
     is obtained, wherein the coefficient K=R2/R1 may be fixed according to needs. 
     Such a divided-down portion V 1  of a base-emitter junction voltage (Vbe Q2 ), as shown in FIGS. 1 and 2, is applied to the base of a first transistor Q 6  of a differential input pair composed of Q 6  and Q 7 , which practically represents a noninverting input of an operational amplifier, configured as a noninverting buffer. The inverting input of the amplifier, represented by the base node of the Q 7  transistor of the differential input pair, is connected to an intermediate node (V 2 ) of a resistive voltage divider R 7 -R 6  of the output voltage produced by the operational amplifier. 
     An analysis of the operation of the circuit of the invention will be rendered more easily by momentarily referring to the partial and simplified circuit diagram of FIG.  2 . 
     Concisely, the transistor pair, Q 6 -Q 7 , and the generator  12  form a differential input stage. The transistor Q 10  and its load, constituted by a diode-configured transistor Q 11  and by a resistance R 5 , constitute an amplifying stage (coupled to R 3A  by a stage including Q 8  and Q 9 ), while the transistor Q 12  constitutes an output stage of the operational amplifier. 
     The amplifier is configured as a noninverting buffer by means of a feedback line constituted by the resistance R7, connected between the output node (Vout) of the amplifier and its inverting input, that is the base node of the transistor Q 7  of the input differential pair, and by the resistance R6 connected between the noninverting input and ground. 
     As already said above, the effectiveness of the voltage reference circuit resides on the fact that the thermal drift of a certain sign of the divided-down portion V 1  of a Vbe voltage, is counterbalanced by a thermal drift of opposite sign of a ΔVbe term (i.e. a voltage difference between two different Vbe voltages), in order to ensure that the resulting sum voltage (V 2 ) has a substantially null temperature coefficient (or thermal drift). 
     To obtain a ΔVbe term to be summed with the divided-down voltage V 1  in order to obtain a resulting sum voltage that is temperature stable, the circuit of FIG. 1 advantageously uses an intrinsic offset voltage of the input pair of transistors Q 6  and Q 7  that form the input differential stage of the operational amplifier. A certain intrinsic offset voltage may be created by appropriately making the two transistors Q 6  and Q 7  that form the input differential pair with different emitter areas. Moreover, the offset voltage is controlled through a dedicated control loop of the bias current that is forced through the input pair of transistors. 
     By referring to the functional diagram of FIG. 2, such a control loop (local feedback) of the bias current forced through the input pair of transistors Q 6  and Q 7  is implemented by the transistors Q 8  and Q 9 , by the respective current generators  13  and  14  and by the resistances R3A and R3B. 
     By assuming negligible (in first approximation) the base current absorbed by the transistor Q 10  and, for example, realizing Q 8  and Q 9  with identical emitter areas and forcing through Q 8  and Q 9  an identical current by the use of identical generators I 3  and I 4 , each capable of generating a current I, the transistors Q 8  and Q 9  will assume an identical Vbe. This, coupled with the fact that the respective bases are connected in common, implies that the emitter voltage of Q 8  is identical to the emitter voltage of Q 9 . This in turn permits to establish a certain current Ib through R 3 B and a certain current Ia through R 3 A, which will have the same ratio (i.e. 1:2) of the value of the resistances R 3 B and R 3 A. As may be observed, the current Ia that flows through R 3 A contains also a contribution coming from the collector of Q 6 . 
     Moreover, by assuming that the current generator  12  of the input differential stage generates a current nI, it is evident that the control loop fixes a certain collector current of the input transistor Q 6  and, as a consequence, the collector current of the other transistor Q 7  of the input differential pair will also be fixed by the local feedback loop, at the value given by the following expression: 
     
       
           Ic   Q7   =nI−I= ( n− 1) I   (1) 
       
     
     By applying Kirchoff&#39;s law, 
     
       
           V   2 = V   1 + Vbe   Q6   −Vbe   Q7   (2) 
       
     
     
       
         
           
             
               
                 
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     It may be observed from the above indicated expressions, that the difference between the respective Vbe voltages of the transistors Q 6  and Q 7  may, in function of the ratio between their respective emitter areas, Ae Q7 /Ae Q6 , assume a temperature coefficient that can be either negative or positive and suitable for compensating the temperature coefficient of a certain sign possessed by the divided-down voltage V 1 . 
     In the depicted example, the divided-down voltage V 1  has a negative temperature coefficient and therefore the intrinsic offset voltage of the differential pair Q 6 -Q 7  must have a positive temperature coefficient. This is achieved by making the transistor Q 7  with an emitter area that is sufficiently larger than the emitter area of Q 6 . Moreover, it is clear that by adjusting the emitter area ratio of the transistors Q 6  and Q 7  and/or the ratio between R 3  and R 2 , a stabilized voltage V 2  may be obtained such that: δV 2 /δT=0. 
     In the circuit diagram of FIG. 1, Q 13 , Q 14 , Q 15 , RE Q1   3  and R 8  constitute a circuit that, through the local feedback, is capable of configuring substantially as a diode the transistor Q 8 , which, together with Q 9 , “reads” the differential stage Q 6 -Q 7 . The signal amplified by Q 10  is transferred through the current mirror Q 11  and Q 12  to the output node Vout, and the resistances R 7  and R 6  close the general feedback loop, by feeding back the V 2 , voltage present on the intermediate node to the base of Q 7  of the input differential stage. (The ratio of R 7 /R 6  is selected to get the desired output voltage V out =V 2 ·R7/R6.) 
     EXAMPLE 
     By assuming a Vbe Q2 =600 mV, with a temperature coefficient of δVbe Q2 /δT=−2 mV/° C., and a partition ratio given by R 2 /R 1 =0.1, a divided-down voltage is obtained that is given by: V 1 =Vbe Q2 R 2 /R 1 =60 mV, having a thermal coefficient of: δV 1 /δT=−2 mV/° C. By assuming n=2, Ae Q7 =10 and Ae Q6 =1, the following is obtained: 
     
       
           ΔVbe=Vbe   Q6   −Vbe   Q7 =60 mV 
       
     
     and 
     
       
           δΔVbe/δT=+ 0.2 mV/° C. 
       
     
     Therefore, the circuit is capable of generating a stabilized voltage: V 2 =120 mV, with δV 2 /δT≈0. 
     In this example, the voltage drop across R 3 A and R 3 B must be maintained equal to or lower than about 200 mV, in order to ensure that the differential pair of transistors Q 6 -Q 7  may function correctly without saturating. 
     The characteristic of a circuit made in accordance with the present invention is shown by the stabilized voltage V 2  versus temperature curve of FIG.  3 . In such an embodiment, without any correction stages, the output voltage V 2  has a temperature coefficient that can be calculated as: 
     
       
           δV   2 /δ° C.=−0.0833 mV/° C. 
       
     
     As will be recognized by those skilled in the art, the innovative concepts described in the present application can be modified and varied over a tremendous range of applications, and accordingly the scope of patented subject matter is not limited by any of the specific exemplary teachings given. For example, as will be obvious to those of ordinary skill in the art, other circuit elements can be added to, or substituted into, the specific circuit topologies shown. For another example, the circuit of the operational amplifier may be realized in a form different from the one depicted in the figures and described above. In particular, stages for correcting the “curvature” of the bandgap characteristic may be added by employing a correction technique similar to the one described in the Gunawan et al. article cited above.