Abstract:
A method and apparatus are disclosed for shaping current in an electronically commutated motor (ECM), the method comprising: acquiring a set of terminal voltage measurements; estimating an instantaneous rotor angle from the terminal voltage measurements; generating a set of phase current commands from the instantaneous rotor angle and from a torque reference signal, each of the phase current commands comprising at least one ripple compensation command segment and at least one zero current command segment; and converting the set of phase current commands to a set of current reference signals.

Description:
BACKGROUND OF INVENTION 
     This invention relates generally to the field of electric motors and particularly to the field of controlling electronically commutated motors. 
     With the advent of reliable, low cost power electronics, an application that once used a brush-type, DC (direct current) motor, is now likely to employ a smaller, safer, cleaner running, longer-lived, and less expensive electronically commutated motor (henceforth, ECM). 
     The ECM is a permanent magnet electrical machine with the magnet (or magnets) on the rotor and multiple, spatially distributed phase windings on the stator. Current in the windings interacts with the permanent magnetic field to produce the machine&#39;s torque. To maintain a constant torque as the rotor turns, the current distribution in the stator is continually adjusted to maintain a constant spatial relationship with the rotor&#39;s magnetic field. The adjustment in current distribution is accomplished by switching (“commutating”) current among the various stator winding phases. In a brush-type DC motor, where the magnets are stationary and the windings rotate, commutation occurs mechanically through the interaction of brushes with a commutator ring. In an ECM, as the name implies, commutation is effected electronically by controlling the conduction states of a multiplicity of electronic power devices electrically coupling the various stator phase windings to a power bus. A subsystem comprising the power devices and any apparatus required to realize a set of power device conduction states appropriate to a respective set of device control signals is called a “power amplifier”; a subsystem comprising any apparatus used to generate the set of device control signals so as to cause the ECM phase currents to track a set of current reference signals is called a “current controller”; a subsystem comprising any apparatus used to generate the set of current reference signals so as to cause the motor torque to track a torque reference signal is called a “torque controller”; a subsystem comprising a power amplifier, a current controller and a torque controller is called a “motor drive”. 
     ECMs are generally divided into two broad classes, distinguished by the way the stator windings are spatially distributed, and named according to the shape of the back EMF (electromotive force) waveforms generated. A motor with a concentrated winding generates a trapezoidally shaped back EMF waveform and is referred to as a “trapezoidal ECM” a motor with a sinusoidally distributed winding is a “sinusoidal ECM.” Conventionally, motor drives used with either class of motor generate phase current waveforms that have a similar shape to and are aligned/with the back EMF waveforms of the respective phases. This current shaping strategy constrains the torque to remain unidirectional. While both motor classes may use the same power amplifier and current controller designs, torque controller designs generally differ between the two. 
     Sinusoidal ECM systems generally employ an explicit rotor angle sensor such as, for example, an optical shaft encoder or a resolver. The torque controller acquires the rotor angle sensor output and generates current reference signals that are sinusoidal functions of the measured angle. 
     In contrast, trapezoidal ECM systems often employ an implicit rotor angle sensing technique. Because the phase current is intermittently conducted, trapezoidal systems can achieve an angle measurement by sensing the back EMF at the terminals of non-conducting phases and generating a signal whenever a prescribed voltage threshold is crossed. These threshold crossing signals initiate step changes in the current reference signals for the appropriate phases. 
     In general, compared to a sinusoidal ECM system, a comparably sized trapezoidal system has the advantage of being less expensive owing to the absence of an explicit angle sensor, the relative simplicity of the torque control algorithm, and the relatively simple construction of the concentrated winding motor. A trapezoidal ECM system has the disadvantage, however, of producing significantly higher levels of undesirable torque ripple. 
     Those applications where cost is the primary concern and torque ripple is a secondary consideration have been traditionally well served by trapezoidal systems. In contrast, those applications requiring lower torque ripple have been traditionally satisfied by more expensive solutions using sinusoidal ECMs. Applications where the torque ripple of a conventional trapezoidal ECM system would be unacceptable, yet the cost of a conventional sinusoidal ECM system would be prohibitive are not well served by either solution. One such application, for example, would be a variable speed blower for high efficiency heating, ventilation, and air conditioning in commercial and domestic buildings. An opportunity exists, therefore, to address these orphan, low ripple, low cost applications. 
     SUMMARY OF THE INVENTION 
     By combining conventional trapezoidal motor, power amplifier, and current controller designs with a novel torque controller design, embodiments of the present invention provide a system with a torque ripple performance approaching that of a sinusoidal system at a cost approaching that of a trapezoidal system. 
     A method and apparatus are disclosed for shaping current in an electronically commutated motor (ECM), the method comprising: acquiring a set of terminal voltage measurements; estimating an instantaneous rotor angle from the terminal voltage measurements; generating a set of phase current commands from the instantaneous rotor angle and from a torque reference signal, each of the phase current commands comprising at least one ripple compensation command segment and at least one zero current command segment; and converting the set of phase current commands to a set of current reference signals. 
    
    
     BRIEF DESCRIPTION OF DRAWINGS 
     These and other features, aspects, and advantages of the present invention will become better understood when the following detailed description is read with reference to the accompanying drawings in which like characters represent like parts throughout the drawings, wherein: 
     FIG. 1 is a block diagram illustrating a trapezoidal ECM coupled to a motor drive in accordance with one embodiment of the present invention. 
     FIG. 2 is a block diagram of one example embodiment of a torque controller for use in the embodiment of FIG.  1 . 
     FIG. 3 is a block diagram of one example embodiment of a current command generator for use in the embodiment of FIG.  1 . 
     FIG. 4 is a block diagram of one example of a rotor angle estimator for use in the embodiment of FIG.  2 . 
     FIG. 5 is a block diagram of another example of a torque controller for use in the embodiment of FIG.  1 . 
     FIG. 6 is a block diagram of another example of a torque controller for use in the embodiment of FIG.  1 . 
     FIG. 7 is a set of graphs illustrating back EMF waveforms as a function of rotor angle for each phase of an ideal three-phase trapezoidal ECM. 
     FIG. 8 is a set of graphs illustrating conventional torque waveforms for individual phase torques and total torque as functions of rotor angle for an ideal three-phase trapezoidal ECM. 
     FIG. 9 is a set of graphs illustrating a set of modified current waveforms for a three-phase trapezoidal ECM in accordance with an embodiment of the present invention. 
     FIG. 10 is a set of graphs illustrating a set of modified torque waveforms for the individual phase torques and the total torque as functions of rotor angle in accordance with an embodiment of the present invention. 
     FIG. 11 is a set of graphs illustrating a set of six-step commands which, when applied to a current controller, produce phase current waveforms with approximately the same shape as the back EMF waveforms depicted in FIG.  7 . 
     FIG. 12 is a set of graphs illustrating a set of phase current commands, each of the phase current commands comprising at least one ripple compensation command segment and at least one zero current command segment, which, when applied to a current controller, produce modified current waveforms approximating those of FIG.  9 . 
     FIG. 13 is a schematic diagram illustrating a conventional, four-leg power amplifier. 
    
    
     DETAILED DESCRIPTION 
     FIG. 1 is a block diagram illustrating a trapezoidal ECM  160  connected to a motor drive in accordance with one embodiment of the present invention wherein the motor drive comprises a torque controller  110 , a current controller  130 , and a power amplifier  150 . As discussed above, trapezoidal ECM  160  has a plurality of stator phase windings. The ends of the stator windings are electrically coupled to terminals on the body of trapezoidal ECM  160  with pairs of such terminals defining motor phases. As FIG. 13 illustrates, power amplifier  150  comprises a collection of electronic power devices  1330  coupling the various motor phases to a power supply bus  1310  and to a power return bus  1320 . In the embodiment of FIG. 13, four phase legs are shown, allowing independent control of all three motor phase currents. In response to a set of device control signals  140  (FIG.  1 ), electrical power is conducted to the appropriate motor phase or phases; power amplifier  150  thus further comprises any electronics required to receive such device control signals  140  and realize the appropriate power device conduction states. 
     Device control signals  140  are generated by current controller  130  whose function is to cause each of ECM phase currents  180  to track a respective one of a set of current reference signals  120 , there being a one-to-one correspondence between the set of current reference signals  120  and the set of ECM phase currents  180 . Typically, as shown, current controller  130  uses feedback from ECM phase currents  180  to accomplish its task. 
     Current reference signals  120  are generated by torque controller  110 . Torque controller  110  receives a torque reference signal  100  which serves as a request for a desired level of total torque in trapezoidal ECM  160 ; torque controller  110  also has access to ECM terminal voltages  170 . The task of torque controller  110  is to determine how the desired total torque is to be distributed among the various motor phases and to generate a set of current reference signals  120 , one signal for each motor phase, so that each phase realizes an appropriate phase torque. The phase torques sum to produce total motor torque. 
     By way of example, but not limitation, an ideal three-phase trapezoidal ECM is considered (FIG. 7) with the phases labeled A, B, and C. Each cycle of a phase A back EMF waveform  701  for trapezoidal ECM  160  comprises: a first ramp EMF segment  700 ; followed by a first constant EMF segment  710 ; followed by a second ramp EMF segment  720 ; followed by a second constant EMF segment  730 ; followed by a third ramp EMF segment  740  ; followed by a third constant EMF segment  730 ; followed by a fourth ramp EMF segment  760 ; followed by a fourth constant EMF segment  770 . The back EMF waveforms for the remaining phases have the same shape as phase A back EMF waveform  701 , with phase B back EMF waveform  702  lagging phase A back EMF waveform  701  by 120 degrees, and phase C back EMF waveform  703  lagging phase B back EMF waveform  702  by 120 degrees. The trapezoidal shape of the non-zero segments of the back EMF waveform gives trapezoidal ECM  160  its name. 
     A conventional approach to the design of a torque controller  110  (FIG. 1) for a three-phase trapezoidal ECM  160  generates current reference signals  120  in the form of six-step commands  1101 ,  1102 ,  1103  (FIG.  11 ), so called because among the three phases there are six non-zero current reference signal steps per rotor cycle. Because the phase windings have a non-zero inductance and because there is only finite voltage available to drive the phase currents, when applied to the current controller these six-step commands  1101 ,  1102 ,  1103  produce phase current waveforms with trapezoidal shapes similar to the back EMF waveforms  701 ,  702 ,  703  (FIG. 7) of the respective phases. For simplicity of presentation, it is herein assumed those current waveforms are identical to the back EMF waveforms  701 ,  702 ,  703 . Since the phase torques obey the equations: 
     
       
           T   A   =E   A   i   A /ω 
       
     
     
       
           T   B   =E   B   i   B /ω, 
       
     
     
       
           T   C   =E   C   i   C /ω 
       
     
     where 
     T A ,T B ,T C    
     are the phase torques in newton-meters, 
     E A , E B , E C    
     are the line-to-neutral phase back EMFs in volts 
     i A ,i B ,i C    
     are the phase currents in amperes, and 
     ω 
     is the rotor angular speed in radians per second, this conventional approach produces conventional phase torque waveforms  801 ,  802 ,  803  (FIG. 8) that are proportional to the squares of the respective back EMF waveforms  701 ,  702 ,  703 . Conventional phase torque waveforms  801 ,  802 ,  803  are, therefore, piecewise constant, corresponding to the constant segments of the back EMF, and piecewise parabolic, corresponding to the ramp segments. Thus, first parabolic torque segment  800  corresponds to first ramp EMF segment  700 , first constant torque segment  810  corresponds to first constant EMF segment  810 , and so forth with torque segments  820 ,  830 ,  840 ,  850 ,  860 , and  870  corresponding to EMF segments  720 ,  730 ,  740 ,  750 ,  760 , and  770 , respectively. 
     Over a range of rotor angles where the phase torque segments for the three phases are all constant, the respective portion of the total motor torque waveform  804  is also constant. For example, a phase A constant torque segment  810 , a phase B constant torque segment  885 , and a phase C constant torque segment  893  sum to produce a total motor torque constant segment  895 . When two of the phase torque segments are parabolic, a torque ripple segment is present in total motor torque waveform  804 . For example, phase A constant torque segment  810 , a phase B parabolic torque segment  880 , and a phase C parabolic torque segment  890  sum to produce a first total motor torque ripple segment  897 . All of the torque ripple segments in total motor torque waveform  804  are formed in a similar fashion: two commutating segments contribute parabolic phase torque segments while a non-commutating segment contributes a constant phase torque segment. As used herein, a segment is said to be “commutating” if it includes at least one zero value and at least one non-zero value; a segment is said to be “non-commutating” if it is not commutating. 
     It is possible to reduce the ripple in total motor torque waveform  804  induced by the commutating segments by modifying the non-commutating segments of the phase current waveforms. By way of example, but not limitation, subtracting first total motor torque ripple segment  897  from a constant torque equal to the torque of total motor torque constant segment  895 , dividing the resulting difference by a constant EMF equal to the EMF of first constant EMF segment  710  (FIG.  7 ), then adding the resulting quotient to the conventional current corresponding to first constant EMF segment  710  yields a phase A modified current waveform  901  (FIG. 9) with a phase A ripple compensation current segment  997 . This phase A modified current waveform  901  produces a phase A modified phase torque waveform  1001  (FIG. 10) including a phase A ripple compensation torque segment  1097  which, when added to phase B parabolic torque segment  880  and to phase C parabolic torque segment  890 , produces a constant segment of a modified total motor torque waveform  1004 . 
     Similar procedures starting with a second total motor torque ripple segment  898  and a third total motor torque ripple segment  899  yield a phase C ripple compensation current segment  998  and a phase B ripple compensation current segment  999 , respectively, which produce a phase C ripple compensation torque segment  1098  and a phase B ripple compensation torque segment  1099 , respectively. Applying this procedure to the remainder of the phase current waveforms produces a set of modified phase current waveforms  901 ,  902 ,  903  which yields a set of modified torque waveforms  1001 ,  1002 ,  1003  which sum to produce modified total motor torque waveform  1004 . Compared to conventional total motor torque waveform  804 , modified total motor torque waveform  1004  shows a reduction in torque ripple. 
     Each of the modified phase current waveforms  901 ,  902 ,  903  (FIG. 9) has two salient features: first, each comprises at least one ripple compensation current segment (for example, phase A ripple compensation current segment  997 ); second, each comprises at least one zero current segment (for example, a phase A zero current segment  930 ). As defined herein, a “ripple compensation current segment” is any portion of a phase current waveform whose purpose is to reduce the total motor torque ripple induced by the other phases; a “zero current segment” is any portion of a phase current waveform wherein the current is zero amperes for at least about 10 microseconds. The purpose of the zero current segments is to allow a back EMF sensing technique to be used for estimating the rotor angle without the use of an explicit rotor angle sensor. 
     An idealized set of current reference signals  120  (FIG. 1) for producing such a set of modified phase current waveforms  901 ,  902 ,  903  (FIG. 9) when applied to a current controller  130  is derived from a set of exemplary phase current commands  1201 ,  1202 ,  1203  (FIG.  12 ), each of which comprises: at least one ripple compensation command segment intended to produce at least one ripple compensation current segment; and at least one zero current command segment intended to produce at least one zero current segment. For example, phase A exemplary phase current command  1201  comprises: a phase A ripple compensation command segment  1297 , intended to produce phase A ripple compensation current segment  997 , and a phase A zero current command segment  1230 , intended to produce phase A zero current segment  930 . As defined herein, a “ripple compensation command segment” is any portion of a phase current command waveform whose purpose is to produce a ripple compensation current segment; a “zero current command segment” is any portion of a phase current command waveform whose purpose is to produce a zero current segment. 
     In one embodiment of the invention, torque controller  110  (FIG. 2) comprises: an ECM interface  200 , a timer  270 , a rotor angle estimator  220 , a current command generator  240 , and a current controller interface  260 . ECM interface  200  is coupled to the terminals of trapezoidal ECM  160  and provides a set of terminal voltage measurements  210  in a format compatible with rotor angle estimator  220 . An instantaneous timer value  280  is maintained by timer  270  which serves as a real-time clock for the system. Rotor angle estimator  220  uses a back EMF threshold crossing sensing technique to compute an instantaneous rotor angle  230  from terminal voltage measurements  210  and from instantaneous timer value  280 ; in some embodiments of the invention, instantaneous rotor angle  230  is the angle directly corresponding to a back EMF threshold crossing event; in other embodiments, instantaneous rotor angle  230  is extrapolated from past back EMF threshold crossing events to provide finer angular resolution. Current command generator  240  computes a set of phase current commands  250  from instantaneous rotor angle  230  and torque reference signal  100 . Each of the phase current commands  250  comprises at least one ripple compensation command segment  1297 ,  1298 ,  1299  (FIG. 12) and at least one zero current command segment  1230 ,  1231 ,  1232 . Current controller interface  260  converts the phase current commands  250  to the set of current reference signals  120  in a format compatible with current controller  130 . 
     In some embodiments of the invention, phase current commands  250  are also a function of instantaneous timer value  280 . For example, in variable speed applications, while the leading edge of the ripple compensation command segment  1297 ,  1298 ,  1299  is initiated at a specified instantaneous rotor angle  230 , it is sometimes desirable for the ripple compensation command segment  1297 ,  1298 ,  1299  to have a calculated time duration rather than a pre-specified angular duration. In such embodiments, the trailing edge of the ripple compensation command segment  1297 ,  1298 ,  1299  is initiated at a calculated instantaneous timer value  280 . Input quantities to the calculated time duration computation may comprise, but are not limited to, torque reference signal  100 , the maximum available ECM terminal voltage  170 , and the electrical impedance of the phase windings of trapezoidal ECM  160 . 
     In a more specific embodiment of the invention, current command generator  240  (FIG. 3) comprises: a unit six-step command generator  300 , a multiplier  320 , a command segment generator  350 , and an adder  340 . Unit six-step command generator  300  generates a set of unit six-step commands  310  from instantaneous rotor angle  230 . By way of example, but not limitation, if torque reference signal  100  represents torque in newton-meters, then unit six-step commands  310  have the same shape as the six step commands  1101 ,  1102 ,  1103  (FIG.  11 ), only unit six-step commands  310  are scaled to produce a modified total motor torque  1004  (FIG. 10) of one newton-meter. Multiplying these unit six-step commands  310  by torque reference signal  100  yields a set of scaled six-step commands  330  capable of producing total motor torque  804  (FIG. 8) with constant segments at the desired torque level. Command segment generator  350  computes a set of ripple compensation command segments  360  from instantaneous rotor angle  230  and torque reference signal  100 . As discussed above, in some embodiments of the invention, ripple compensation command segments  360  are also functions of instantaneous timer value  280 . Adding respective ones of ripple compensation command segments  360  to scaled six-step commands  330  yields phase current commands  250  which deliver a modified total motor torque  1004  at the desired torque level. 
     In another embodiment of the invention, rotor angle estimator  220  (FIG. 4) comprises: an event recorder  430 , a function constructor  470 , and a function evaluator  495 . An angle estimation function  490  is evaluated at instantaneous timer value  280  by function evaluator  495  to yield instantaneous rotor angle  230 . Whenever any of terminal voltage measurements  210  crosses a voltage threshold  400 , event recorder  430  includes instantaneous timer value  280  in a set of recorded timer values  460 . Function constructor  470  then constructs a new angle estimation function  490  from the set of recorded timer values  460  and from a set of respective, pre-stored rotor angles  480 . 
     In another embodiment of the invention, torque controller  110  (FIG. 5) comprises: an analog-to-digital converter  500 , a current controller interface  260 , and a digital signal processor  520 . Analog-to-digital converter  500  serves as an ECM interface  200  and provides a set of terminal voltage measurements  210  in a format compatible with digital signal processor  520 . Current controller interface  260  converts a set of phase current commands  250 , generated by digital signal processor  520 , into a set of current reference signals  120  compatible with current controller  130 . Digital signal processor  520  implements rotor angle estimator  220  and current command generator  240  in software. As defined herein, the term “digital signal processor” includes, but is not limited to, single-chip digital signal processors (i.e., DSP chips), microprocessors, microcontrollers, bit-slice processors, and application specific integrated circuits (ASICs). 
     In still another embodiment of the invention (FIG.  6 ), analog-to-digital converter  500  is replaced by an electronic threshold crossing detector  600 . Electronic threshold crossing detector  600  generates a threshold crossing signal  610  whenever ECM terminal voltages  170  cross a voltage threshold  400 . Thus, the threshold detection function of event recorder  430  is implemented in hardware. Digital signal processor  520  responds to threshold crossing signal  610  by including instantaneous timer value  280  in a set of recorded timer values  460  (FIG. 4) and otherwise completing the functions of rotor angle estimator  220  and current command generator  240 . 
     While only certain features of the invention have been illustrated and described herein, many modifications and changes will occur to those skilled in the art. It is, therefore, to be understood that the appended claims are intended to cover all such modifications and changes as fall within the true spirit of the invention.