Abstract:
A source ( 20 ) provides an input signal (S 1 ) to be phase shifted and a combining circuit ( 24 ) concurrently combines first (A), second (B) and third (C) intermediate signals derived from the input signal (S 1 ), and having differing phase shifts (0, −45, +135 deg), to form a phase shifted output signal (S 2 ). A first amplitude controller ( 34, 38, 30, 32 ), responsive to a phase control signal (S 3 ) supplied thereto, varies the amplitudes of the second (B) and third (C) intermediate signals in opposite directions ( 38 ) for controlling the phase of the phase shifted output signal. Additionally, a further amplitude controller ( 40, 42 ) is provided for reducing a tendency for variations in the phase shift control signal (S 3 ) to alter the amplitude of alternating current (FIG.  5 ) and direct current (FIG.  6 ) components of the phase shifted output signal (S 2 ).

Description:
FIELD OF THE INVENTION 
     This invention relates to phase shift apparatus and particularly to variable phase shifters of a type in which plural phase shifted components of an input signal are formed and combined in proportions determined by a phase shift control signal to provide a phase shifted output signal. 
     BACKGROUND OF THE INVENTION 
     Variable phase shift apparatus are useful, for example, as phase control elements in voltage controlled oscillators, tuning circuits, filters and in similar applications where controlled phase shift is needed. For example, in U.S. Pat. No. 4,020,500 entitled CONTROLLED OSCILLATOR which issued Apr. 26, 1977 to L. A. Harwood, there is described a voltage controlled crystal oscillator in which phase shift for controlling the oscillator frequency is provided by vector summation. Specifically, an oscillator signal is phase shifted by +90 degrees by means of a two-pole L-C low pass filter and combined with the original signal to produce a resultant vector lying between 0 degrees and a leading angle (e.g., +45 degrees) with the angle being controllable by varying the amplitude of the +90 degree (quadrature) signal. For providing lagging phase shifts (e.g., from 0 degrees to about −45 degrees) the polarity of the quadrature vector is inverted to be −90 degrees rather than +90 degrees and the inverted quadrature vector is added to the reference or input signal. Here the resultant vector lies between 0 degrees and a lagging angle between 0 degrees and about −45 degrees with the angle being controllable by varying the amplitude of the −90 degree vector. 
     A phase shift circuit, in an oscillator application, which avoids the need for quadrature phase shifting elements (e.g., 90 degree two-pole filtering) is described by Paul D. Filliman in his U.S. Pat. No. 4,797,634 entitled CONTROLLED OSCILLATOR which issued Jan. 10, 1989. Unlike the Harwood arrangement, the Filliman phase shifter combines three vectors at a time rather than two. The vector angles used are −45 degrees, 0 degrees and +135 degrees with respect to the input signal and the mixture of these three vectors provides all phase shifts, both positive (leading) and negative (lagging). Also, since the +135 degree vector is produced from the −45 degree vector by inversion, and since the −45 degree shift may be provided by a one-pole RC filter, the entire phase shift circuit may be readily constructed in integrated circuit form in applications such as providing phase control of color oscillators in television related products. 
     FIG. 1 herein presents a simplified block diagram of the Filliman oscillator. As shown, a phase shifter  10  is provided with two feedback paths comprising (i) a DC path ( 12 ) via a resistor ( 14 ) which regulates the DC operating point of the oscillator by means of negative feedback and (ii) an AC feedback path ( 16 ) through a resonator ( 18 , e.g., a crystal) which provides positive feedback with a gain of unity for causing oscillations to occur. 
     Phase shifter  10 , as illustrated in FIG. 1, is non-inverting whereas oscillator applications require inversion to provide the proper negative DC feedback for bias stabilization and positive AC feedback signal at the nominal resonator frequency, Fr, for causing oscillations to occur. The inversion of the feedback signal for oscillator applications is provided in FIG. 1 by an inverter  35  at the phase shifter output  22 . Optionally, inversion may be applied at various places. For example, the inverter  35  may just as well be placed at the input  20  of the phase shifter  10  or within the phase shifter. Also, inversion may be provided by simply using the inverting output of a differential output amplifier in the phase shift network  10  or by using an inverting form of summing amplifier  24  at the phase shifter output. What is of importance to the present invention is not whether the shifter inverts or not, but how the phase shifting is accomplished as will be described. 
     Phase shifter  10  includes an input terminal  20  for receiving the oscillator input signal S 1  to be phase shifted and an output terminal  22  for providing a phase shifted output signal S 2 . The input signal S 1  separated into three “intermediate” signals or “vector” components “A”, “B” and “C” which are summed via a summing circuit  24  to provide a phase controlled output signal S 2  at the network  10  output terminal  22 . Hereinafter, the terms “intermediate” signals or components may be used interchangeably with the term “vector” signals or components. 
     The processing of the input signal S 1  to form the intermediate signals or vector components “A”, “B” and “C” is provided by a constant gain amplifier  28 , two variable gain amplifiers  30  and  32  (the latter of which is inverting) and a 45 degree phase lag network  26 . Vector “A” in FIG. 1 has a phase angle of zero degrees with respect to the input signal S 1  and is produced by applying the input signal to the summing circuit  24  via a non-inverting fixed gain amplifier  28 . 
     Vector “B” has a phase angle of −45 degrees relative to the input signal S 1  and is produced by phase shifting the input signal by 45 degrees (lagging) at the nominal frequency of the resonator  18  in the phase shift network  26  (e.g., a low pass filter). The phase shifted signal “B” is then applied to summing circuit  24  via a variable gain non-inverting amplifier  30 . 
     Vector “C” has a phase angle of +135 degrees relative to the input signal S 1  and is produced by inverting the output of the 45 degree phase shift network  26  with a controllable gain inverting amplifier  32  for application to summing circuit  24 . A phase shift control signal  53  is applied directly to the gain control input of amplifier  30  and is inverted by means of an inverter  38  for application to the gain control input of inverting amplifier  32 . 
     In operation, due to the action of the inverter  38 , an increase in control signal S 3  at terminal  22  will cause an increase in the amplitude of the vector “B” and concurrently will cause a decrease in the amplitude of the vector “C”. This will produce a lagging phase for the output signal S 2  relative to the input signal S 1 . Conversely, a decrease in the control signal S 3  will cause a decrease in the amplitude of vector “B” and an increase in the amplitude of vector “C”. This will produce a leading phase for the output signal S 2  relative to the input signal S 1 . When changing phase angles, the amplitude of vector “A” is held constant. It is the variation of the amplitudes of vectors “B” and “C” which controls the net phase shift of the network and thus the frequency of the oscillator. 
     SUMMARY OF THE INVENTION 
     It has been found that under certain special circumstances (e.g., improper amplifier gains or gain variations) the Filliman type of oscillator, employing a three-vector phase shifter of the type described, may exhibit one, or more, of the following three operating difficulties: (i) stopped oscillations; (ii) overtone oscillations; and (iii) bias instability. 
     The problems noted above have been found to be traceable to the phase shift portion of the oscillator. It has also been found that the problems with the oscillator may be reduced by the expedient of restricting the phase range of the phase shifter. This, however, is not a completely satisfactory solution since it limits the oscillator frequency range and thus limits the range of useful applications for the oscillator. 
     Phase shift apparatus, embodying the invention, includes a source providing an input signal to be phase shifted and a combining circuit for combining first, second and third intermediate signals that are derived from the input signal, and have differing phase shifts, to form a phase shifted output signal. A first amplitude control circuit, responsive to a phase control signal supplied thereto, varies the amplitudes of the second and third intermediate signals in opposite directions for controlling the phase of the phase shifted output signal. Additionally, means are provided for reducing a tendency for variations in said phase shift control signal to alter the amplitude of the phase shifted output signal. 
    
    
     BRIEF DESCRIPTION OF THE DRAWING 
     The foregoing features of the invention are illustrated in the accompanying drawing, wherein like elements are denoted by like reference designators, and in which: 
     FIG. 1 is a simplified block diagram of a prior art oscillator employing a phase shift network of a three vector summation type; 
     FIGS. 2A-2D are vector diagrams illustrating certain aspects of normal operation of the phase shift network of FIG. 1; 
     FIGS. 3A-3D are “pseudo” vector diagrams illustrating certain aspects of abnormal operation which has been found to occur under certain circumstances in the phase shift network of FIG. 1; 
     FIG. 4 is a block diagram of a phase shift network embodying the invention; 
     FIGS. 5A-5D are vector diagrams illustrating normal operation of the phase shifter of FIG. 4; 
     FIGS. 6A-6D are “pseudo” vector diagrams illustrating certain aspects of operation of the phase shifter of FIG. 4; 
     FIGS. 7,  8 A and  8 B are circuit diagrams illustrating suitable minimum gain limiting circuits suitable for use in the example of FIG. 4; and 
     FIGS. 9,  10  and  11  are block diagrams illustrating modifications of the phase shifter of FIG.  4 . 
    
    
     DETAILED DESCRIPTION 
     Before considering the details of the present invention, it is helpful to consider why the prior art three vector phase shifter of Filliman is subject to stopped oscillations, overtone oscillations and bias instability under certain circumstances and why restricting the phase control range overcomes these difficulties. 
     First, as to the stopped oscillations, if the open-loop gain ever drops below unity, the conditions for oscillation are no longer met. This can happen if the voltage-controlled phase shift circuit exhibits a significant decrease in gain at the fundamental frequency of the resonator for some values of control voltage. This problem is illustrated by FIGS. 2A and 2C. FIG. 2A is a vector diagram illustrating the three basic vectors in the phase shift circuit of FIG. 1, namely, “A” (at 0 degrees), “B” (at −45 degrees) and “C” at +135 degrees. All are assumed to have similar amplitudes. FIG. 2C illustrates the resultant phase shift (vector “R”) for a condition in which the phase angle control signal is at a minimum value (e.g., zero). In this case vectors “A” and “C” when summed in summing circuit  24  produce a resultant vector “R” that is much smaller than either vector “A” or vector “C”. From this it is apparent that, at the minimum phase control signal value (S 3 =0), the gain of the phase shift circuit is substantially less than unity and so oscillations may cease. Accordingly, for this case one would like to increase the phase shifter gain. 
     Second, regarding the overtone oscillations, if the open-loop gain ever exceeds unity at an out-of-band frequency, a condition for spurious oscillation is fulfilled. This typically happens at the third overtone of the crystal, which acts like a bandpass filter at the fundamental frequency and at odd harmonics of the fundamental. A low-pass filter is normally incorporated in crystal networks (resonator) to curb this problem but excess gain in the voltage-controlled phase shift circuit as a function of control voltage can still result in overtone oscillation. FIG. 2B illustrates this excess gain condition. In FIG. 2B the phase shift control signal S 3  is shown at a maximum value. In this case, because of the inversion in inverter  32 , the vector “C” is zero and the vectors “A” and “B” are both full valued. Their vector sum or the resultant output signal “R” produced by summing circuit  24  is much larger than either of the individual vectors thus indicating a gain increase. If, in oscillator applications, the gain increase is sufficiently large, as noted above, spurious oscillations may occur. Thus for this case, one would wish to decrease the phase shifter gain. 
     In the above two examples it is seen that for sustained oscillations one would like to increase the gain of the phase shift network at minimum values of the phase shift control signal S 3 . However, for reduced spurious signals one would like to decrease the phase shifter gain at maximum values of the control signal S 3 . These differing gain requirements are illustrated in the composite vector diagram of FIG. 2D which presents the resultant vectors for minimum, maximum and midrange values of the control signal S 3 . To solve this problem one might be tempted to merely vary the gain of the buffer amplifier  35  as a function of the control signal S 3 . Such a hypothetical solution, however, will not work. 
     In accordance with an aspect of the invention, the reason that it is not possible to correct the problems of stopped oscillations and overtone oscillations by merely varying the amplitude of the resultant vector “R”, is because of a third gain problem concerning the circuit operation at very low frequencies. By “low” frequencies it is meant that the phase shift provided by the 45 degree phase shifter  26  is reduced to a very low value (e.g., a few degrees) or, at zero Hz where there is no phase shift at all provided by low pass filter  26 . 
     Specifically, it has been found that at very low, or zero, frequencies, where filter  26  provides no significant delay and so no appreciable phase shift, that under a certain condition of the control signal S 3  the “polarity” of the resultant vector “R” may be reversed. 
     More specifically, if the “DC” or low frequency gain approaches zero (or worse yet, crosses zero and changes polarity), bias instability can result. Depending on the DC bias method used in the application circuit, changes in DC or low frequency gain can shift the DC operating point of the loop, resulting in stopped oscillation or an unsatisfactory duty cycle or, for the phase inversion case, a latch-up condition in which the DC feedback is positive and the circuit acts like a latch. This problem is illustrated by the vector diagrams of FIGS. 3A-3D. These diagrams are similar to those of FIGS. 2A-2D except that they are made for the condition that the input signal frequency is zero (i.e., it is a direct current, DC). 
     At DC, the low pass filter  26  provides approximately 3 dB higher amplitude than at the nominal (resonator) frequency and there is no associated phase shift. The reason for the 3 dB gain increase is that the low pass filter  26  introduces a 3 dB loss at the resonator frequency to obtain a −45 degrees of phase shift and this loss is corrected by adjustment of the gains of amplifiers  30  and  32  or the weighting of summer  24 . This, at DC the amplitude compensated low pass filter will exhibit a net gain of 3 dB (i.e., a gain of 1.414). Accordingly, the vectors “B” and “C” are larger than vector “A” and are on-axis with respect to vector “A”. This is illustrated by the basic vectors of FIG.  3 A. For both of these reasons (B,C&gt;A &amp; all on axis) the voltage controlled phase shifter exhibits an even higher gain sensitivity (to the control voltage S 3 ) at DC and low frequencies than at the nominal frequency of the resonator  18 . 
     FIG. 3B shows the resultant “R” for the case where the phase control signal S 3  is a maximum. FIG. 3C illustrates the opposite case where S 3  is a minimum (zero). For this case vector “B” is zero and vector “C” is at its maximum. This, however, at DC and low frequencies, is 3 dB greater than vector “A”. As a result, the phase of the resulting vector “R” is reversed 180 degrees. This, in a oscillator application where the DC feedback must be negative, will cause positive feedback and therefore latch-up of the overall circuit stopping the oscillations. This undesirable condition is shown in FIG. 3D which illustrates a composite of the three resultant vectors for minimum, midrange and maximum values of the control signal S 3 . As seen, with the phase control signal S 3  at its minimum value, the phase of the resultant vector “R” is reversed. 
     FIG. 4 is a block diagram of phase shift apparatus embodying the invention in which means are provided for reducing the tendency for variations in the phase shift control voltage S 3  from altering the amplitude of the phase shifted output signal S 2 . This is achieved in a way such that the phase of the phase shifted output signal does not reverse for any value of the phase control signal. 
     The phase shifter  10  in FIG. 4 is similar to the phase shifter of FIG. 1 except that (1) the fixed gain amplifier  28  of FIG. 1 is replaced by a variable gain amplifier  40  in FIG. 4, (2) the gain control for amplifier  40  is obtained from the same bus  52  that controls the gain of inverting amplifier  32  and (3) a minimum gain limiter  42  is connected to the variable gain amplifier  40 . 
     Operation of the phase shifter  10 A is illustrated by FIGS. 5A-5D and  6 A- 6 D. FIG. 5A illustrates the three basic vectors used in the summation which are of equal amplitude and at angles of 0, −45 and +135 degrees. 
     FIG. 5B illustrates the case where the phase control signal S 3  is at a maximum value thus making an increase in vector “B”, a decrease to zero of vector “C” and reducing vector “A” to the minimum value determined by the minimum gain limiter  42 . This produces a lagging resultant vector R which is slightly larger than vector “B”. 
     FIG. 5C illustrates the other extreme of the control signal S 3  when S 3  is set to a minimum value (e.g., zero). In this case the gain of amplifier  30  is reduced to zero so the amplitude of vector “B” is also zero. The amplitude of vector “A” is increased by amplifier  40 . The sum “R” of the vectors “C” and “A” is a leading vector of about the same size as in FIG.  5 B. FIG. 5D is a composite vector diagram illustrating the resultant vectors for minimum, midrange and maximum values of the phase control signal S 3 . As shown, there is little difference in amplitude of the resultant vector “R” over the full range of signal S 3 . 
     From the foregoing, it has been seen that the gain of the overall phase shifter circuit  10 A has been stabilized at the nominal operating frequency where the phase shift of filter  26  is 45 degrees. Consideration will now be given to the amplitude response at DC and low frequencies where the phase shift of filter  26  is negligible. 
     As in the previous discussion, it is assumed that the gains of all vectors are normalized at the nominal frequency of operation of the phase shifter. This would be near the resonator frequency in oscillator applications. This at DC and low frequencies the vectors “B” and “C” at a mid range setting will be larger (by 3 dB) than vector “A”. FIG. 6A illustrates this vector relationship of the “basic” vectors at DC and low frequencies. FIG. 6B illustrates a case where the phase control signal S 3  is set to a maximum value. In this case the amplifier  32  will reduce the magnitude of vector “C” to zero and amplifier  40  will reduce the amplitude of vector “A” to the minimum value determined by the minimum gain limiter  42 . Thus the resultant vector “R” will equal the sum of the minimum “A” value plus the increased amplitude of vector “B”. As illustrated, there is no polarity of the resultant vector for this maximum setting of the phase control signal S 3 . 
     FIG. 6C illustrates a case where the phase control signal S 3  is reduced to its minimum (zero) value. This decreases the gain of amplifier  30  thus making the amplitude of vector “B” equal to zero. Amplifiers  40  and  32  both exhibit increase gains since they receive the inverted (S 3 -bar) phase control signal. The maximum gain of amplifier  40 , however, is selected to be sufficiently larger than that of amplifier  32  to that the vector “A” is larger than vector “C” under the assumed condition. A suitable gain difference is on the order of about 6 dB which allows for some amplifier gain tolerance variations. As a result, the resultant vector “R” is positive (i.e., non-inverted) for this minimum setting of the phase control signal S 3 . FIG. 6D is a vector diagram that is a composite of three values of the phase control signal, namely, minimum, midrange and maximum. Although this diagram shows gain variation at DC and low frequencies, the phase does not invert for any setting of the phase control signal S 3 . 
     In the foregoing example of the invention, a variable gain amplifier  40  was used for varying the amplitude of the intermediate signal (vector “A”) and a minimum gain limiter  42  was connected to the amplifier  40 . FIG. 7 is a detailed circuit diagram of a variable gain differential amplifier suitable for use as amplifier  40  and minimum gain limiter  42 . The amplifier includes a pair of source coupled transistors Q 3  and Q 4  having drain loads R 1  and R 2  coupled to ground and to output terminals  706  and  707 . The quiescent currents of transistors Q 3  and Q 4  are varied to vary the transconductance (gm) of the amplifier thereby producing a variable gain K proportional to the product of the variable transconductance and the load value (RL). 
     For transconductance control, a transistor Q 2  is provided operating at a fixed gate to source bias applied to terminals  701  and  703 . This transistor, in other words, is biased for operation as a fixed current source. Accordingly, the transconductance of the differential amplifier transistors Q 3  and Q 4  can not be less than that determined by the constant current provided by transistor Q 2 . This corresponds to the action of the minimum gain limiter  42  in FIG.  4 . 
     Variable gain control is provided by a transistor Q 1  having a fixed source voltage an a gate voltage that varies with the control signal applied to the control input  702  thus forming a variable current source. Thus as the phase control signal varies the current provided by transistor Q 1  also varies thus controlling the transconductance of transistors Q 3  and Q 4 . When the gain control voltage applied to terminal  702  biases transistor Q 1  off, however, the transconductance does not reduce to zero since transistor Q 2  still supplies a minimum current independent of the current provided by transistor Q 1 . The maximum gain is determined by the sum of the currents provided by both of the current source transistors. 
     There are various other ways of setting a minimum amplitude for the vector “A”. For example, FIG. 9 illustrates a modification of the phase shifter of FIG. 4 in which the minimum gain limiter  42  is placed in the control signal path for the gain control signal (S 3 -bar) of amplifier  40 . For this purpose a clamp circuit may be used. FIGS. 8A and 8B illustrate two suitable clamp circuits. In FIG. 8A the control signal form inverter  38  is applied to an input terminal  801  which is coupled to an output terminal  803  via a diode  802 . Terminal  803  is also coupled to ground via a resistor  804  and a source of reference voltage (illustrated as a battery)  805 . 
     In operation, when the phase control signal is greater than the sum of the battery voltage and the forward drop of the diode  802 , the diode will become conductive and supply a control signal to the variable gain amplifier  40  that varies with the phase control signal. However, at lower voltages the diode will be non-conductive and so the voltage applied to the gain control input of amplifier  40  will be a constant value equal to the reference source  805  voltage. The clamp circuit of FIG. 8B is similar to that of FIG. 8A except that the diode  802  and resistor  804  are interchanged. In operation, for input voltages greater than the sum of the battery and diode voltages, the diode will be turned off and the output voltage to amplifier  40  will follow the control signal at terminal  801 . Conversely, at lower voltages the diode will be forward biased thereby maintaining a constant gain control bias for amplifier  40  and thus establishing a minimum gain. 
     FIG. 10 is a block diagram illustrating a modification of the example of FIG. 4 in which the minimum gain limiter  42  is implemented by placing a by-pass path  600  containing an amplifier  620  around amplifier  40 . This path ensures that even when amplifier  40  is biased to provide a zero output, vector “A” will still have a minimum value determined by the attenuator. FIG. 11 illustrates a modification of the phase shifter of FIG. 4 in which the inverter  38  is moved from phase control signal bus  52  to phase control signal bus  50 . The circuit operation is the same as in FIG. 4 except that the phase shifts in the opposite direction as the phase control signal S 3  varies.