Abstract:
In one embodiment, a pressure sensor assembly for use with an application specific integrated circuit includes a capacitive sensor, a sensor coil within a first sensor compartment and operatively connected to the capacitive sensor to form a sensor L-C tank circuit, a measuring oscillator including a measuring coil located within a second sensor compartment and spaced apart from the sensor coil and a feedback circuit configured to provide a control signal for the measuring oscillator based upon an output of the measuring oscillator, and a low frequency signal source configured to provide a low frequency signal to the measuring oscillator.

Description:
FIELD OF THE INVENTION 
     This invention relates to the field of pressure sensors and more particularly to capacitive pressure sensors. 
     BACKGROUND OF THE INVENTION 
     As systems become more sophisticated, incorporation of increased amounts of data in controlling those systems is useful in maximizing the system performance. Thus, in automotive applications, data associated with the pressure of various media including air, gasoline or transmission oil may be used to optimize engine performance. 
     One practice in engine design directed toward increased fuel efficiencies is to mix engine exhaust with fresh air so as to preheat the incoming air. Obtaining the pressure of the resultant air/exhaust mixture, however, necessitates exposing a sensor element to a significantly more aggressive and hotter environment. Accordingly, contemporary pressure sensors require special protection of the electrical connections on the sensor. 
     One approach to protecting sensitive components of a sensor assembly from harsh environments is to deposit a gel over the sensor and electronics, thereby sealing the vulnerable electrical connections from the harsh media. Other sensors physically separate sensitive electronics from the harsh environment. In these sensors, one of a pair of coils, along with the sensitive electronics, is placed in a protected environment. A second coil an substrate, made of a material that is resistant to the harsh environment, is positioned in the harsh environment and data is passed between the coils. 
     While effective in isolating the sensitive components of the sensor from the harsh environment that is monitored by the sensor assembly, sensors incorporating additional materials introduce additional manufacturing steps. Moreover, the protective materials tend to break down over time and lose the ability to protect the sensor, thereby shortening the useful life of the sensor. 
     Additionally, as a monitored engine continues to be operated, the temperature of the exhaust varies. The change in temperature can affect the sensor assembly characteristics thereby exacerbating sensor system inaccuracies. Some of the temperature related inaccuracies may be mitigated by measuring the temperature and applying a temperature correction. As the number of sensors increases, however, costs associated with the system increase. Additionally, additional space is required for the additional component. 
     Accordingly, a sensor assembly which protects sensitive components from harsh environments without requiring protective materials on sensor components would be advantageous. A sensor assembly that provided temperature data along with pressure data would be further advantageous. A sensor assembly that used temperature data to optimize sensor system accuracy would be further advantageous. 
     SUMMARY OF THE INVENTION 
     In one embodiment, a pressure sensor assembly for use with an application specific integrated circuit includes a capacitive sensor, a sensor coil within a first sensor compartment and operatively connected to the capacitive sensor to form a sensor L-C tank circuit, a measuring oscillator including a measuring coil located within a second sensor compartment and spaced apart from the sensor coil and a feedback circuit configured to provide a control signal for the measuring oscillator based upon an output of the measuring oscillator, and a low frequency signal source configured to provide a low frequency signal to the measuring oscillator. 
     In another embodiment, a method of identifying a pressure of a fluid includes coupling a measuring oscillator which is isolated from the fluid with a sensor component, establishing a circulating current within the sensor component through the coupling, exposing a capacitive sensor of the coupled sensor component to the fluid, establishing a resonant frequency of the sensor component based upon the exposure, varying the frequency of the coupled measuring oscillator, generating a change in voltage for the measuring oscillator with the varied frequency, controlling the frequency of the measuring oscillator based upon the generated change in voltage, and generating a signal associated with the pressure of the fluid based upon the controlled frequency of the measuring oscillator. 
     The above described features and advantages, as well as others, will become more readily apparent to those of ordinary skill in the art by reference to the following detailed description and accompanying drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The present invention may take form in various system components and arrangement of system components. The drawings are only for purposes of illustrating exemplary embodiments and are not to be construed as limiting the invention. 
         FIG. 1  depicts a cross-sectional perspective view of a sensor assembly with sensitive electronics protected from the environment that is monitored in accordance with principles of the invention; 
         FIG. 2  depicts a perspective view of the sensor used in the sensor assembly of  FIG. 1  including a capacitive sensor and a loop which form a tank circuit; 
         FIG. 3  depicts a system diagram of a circuit on the application specific integrated circuit of the sensor assembly of  FIG. 1  that provides automatic tuning of a measuring oscillator to match the resonant frequency of a tank circuit of the sensor of  FIG. 1 ; 
         FIG. 4  depicts the measured amplitude of the measuring oscillator in the system of  FIG. 3  with a fixed gain as the frequency of the measuring circuit is tuned through frequencies including the resonant frequency of the sensor of  FIG. 2  showing a dip that results when the frequency of the measuring oscillator is near the resonant frequency of the sensor tank circuit; 
         FIG. 5  depicts the low frequency signal applied to the measuring oscillator of  FIG. 3  and the changes in the signal behind the demodulation filter which results if the frequency of the measuring oscillator is at the resonant frequency of the sensor tank circuit, if the frequency of the measuring oscillator is below the resonant frequency of the sensor tank circuit, and if the frequency of the measuring oscillator is above the resonant frequency of the sensor tank circuit; 
         FIG. 6  depicts the signal behind the controller output of the system of  FIG. 3  as the frequency of the measuring circuit is modulated with a low frequency signal when the frequency of the measuring oscillator is matched with the resonant frequency of the sensor of  FIG. 2 ; 
         FIG. 7  depicts the signal behind the controller output of the system of  FIG. 3  as the frequency of the measuring circuit is modulated with a low frequency signal when the frequency of the measuring oscillator is below the resonant frequency of the sensor of  FIG. 2 ; 
         FIG. 8  depicts the signal behind the controller output of the system of  FIG. 3  as the frequency of the measuring circuit is modulated with a low frequency signal when the frequency of the measuring oscillator is above the resonant frequency of the sensor of  FIG. 2 ; 
         FIG. 9  depicts a schematic diagram of a circuit on the application specific integrated circuit of the sensor assembly of  FIG. 1  that provides automatic tuning of a measuring oscillator to match the resonant frequency of a tank circuit of the sensor of  FIG. 1  while reducing noise because the circuit topology of  FIG. 9  prevents a frequency dependent gain change; 
         FIG. 10  depicts a top plan view of a switched capacitor that may be used as a variable capacitor in the measuring oscillator of the sensor assembly of  FIG. 1 ; 
         FIG. 11  depicts a schematic diagram of a circuit on the application specific integrated circuit of the sensor assembly of  FIG. 1  with a measuring oscillator including threshold dependent automatic amplitude gain control that provides a system as shown in  FIG. 3  the ability to form a self-tuning system of a measuring oscillator to match the resonant frequency of a tank circuit of the sensor of  FIG. 1  while reducing noise within the system by incorporating a variable current gain operational transconductance amplifier; 
         FIG. 12  depicts a schematic diagram of a circuit on the application specific integrated circuit of the sensor assembly of  FIG. 1  with a single ended feedback circuit with dual inputs that provides a system as shown in  FIG. 3  the ability to form a self-tuning system of a measuring oscillator to match the resonant frequency of a tank circuit of the sensor of  FIG. 1  while reducing noise within the system by incorporating a variable current gain operational transconductance amplifier; 
         FIG. 13  depicts a schematic diagram of an input stage that may be used with the variable current gain operational transconductance amplifier of  FIG. 12 ; 
         FIG. 14  depicts various measuring oscillator frequency response curves without sensor coupling that may be obtained by implementing a trimable capacitor in the circuit of  FIG. 12 ; and 
         FIG. 15  depicts a measuring oscillator frequency response curve that may be obtained by implementing a trimable capacitor of  FIG. 14  in the circuit of  FIG. 12  with a sensor coupled to the measuring oscillator. 
     
    
    
     DESCRIPTION 
     Referring to  FIG. 1 , a sensor assembly  100  includes a housing  102  and a sense port  104 . The sense port  104  includes a coupling portion  106  and a neck  108 . A bore  110  extends from the coupling portion  106  to a sensor compartment  112 . The sensor compartment  112  is isolated from an electronics compartment  114  by a substrate  116  which in this embodiment is integrally formed with the housing  102 . A sensor  118  is positioned within the sensor compartment  112  and an electronic assembly  120  is positioned within the electronics compartment  114 . 
     The sensor  118 , also shown in  FIG. 2 , includes a capacitive pressure sensor  122  and a coil  124 . The coil  124  is connected to the capacitive pressure sensor  122  by electrodes  126  and  128 . The sensor  118  thus forms an LC tank circuit with the capacitive pressure sensor  122  functioning as a capacitor which varies in capacitance as the pressure within the sensor compartment  112  varies while the coil  124  functions as an inductor. 
     The electronic assembly  120  includes a primary coil  130  and an application-specific integrated circuit (ASIC)  132 . The electronic assembly  120  is depicted schematically in  FIG. 3 . In  FIG. 3 , the primary coil  130  is in parallel with a variable capacitor  140  to form a measuring oscillator  142 . The measuring oscillator  142  may be a voltage control oscillator or a digitally controlled oscillator. The output of the measuring oscillator  142  is provided to a demodulator  144 . The demodulator  144  further receives a low frequency signal from a low frequency source  146  through an optional phase corrector  148 . Typically, the phase correction is small and in certain applications the phase corrector  148  may be omitted. 
     The output of the demodulator  144  is provided to an optional filter  150  and the signal from the filter  150  is passed to a proportional-integral-derivative (PID) controller  152 . The PID controller  152  output is provided to a summer  154  which combines the output of the PID controller  152  with the output of the low frequency source  146 . The output of the summer  154  is used to control the variable capacitor  140 . An optional frequency counter  156  is provided to count the cycles of the measuring oscillator  142  and the measured frequency is provided as an output  158 . Alternatively, the output of the PID controller  152  may be used as an output. By way of example, the average capacitor value may be used to ascertain the pressure sensor capacity directly. Increased accuracy may be obtained by providing temperature compensation for the measuring oscillator  142  using this alternative approach. 
     In operation, the electronic assembly  120  is used to identify the resonant frequency of the sensor  118 . To this end, the variable capacitor  140  and the primary coil  130  function as a variable frequency oscillator circuit of the measuring oscillator  142 . The frequency of the oscillation of the measuring oscillator  142  is modified by changing the capacitance of the variable capacitor  140 . In general, as the measuring oscillator  142  oscillates, the primarily coil  130  establishes a magnetic field which loosely couples the primarily coil  130  and the coil  124  in the sensor  118 . As the frequency of the measuring oscillator  142  approaches the resonant frequency of the sensor  118 , a circulating current is developed within the sensor  118  through the established coupling. 
     Because of the circulating current within the coil  124 , an impedance is reflected through the coupling between the primary coil  130  and the coil  124  to the primary coil  130 , causing an increase in loop gain y he PID controller  152  in order to compensate for the energy drain and keep the amplitude of the measuring oscillator  142  stable. At the resonant frequency of the sensor  118 , the circulating current within the coil  124 , and thus the reflected impedance into the measuring oscillator  142 , is maximized. 
       FIG. 4  graphically illustrates the general functioning of the electronic assembly  120 . The line  160  represents the measured voltage in a constant current device as frequency of the measuring oscillator  142  is modified from a lower frequency  162  to a higher frequency  164  by changing the capacitance of the variable capacitor  140 . As the frequency of the measuring oscillator  142  increases from the frequency  162  to the frequency  164 , a dip  166  is observed in the voltage as a result of the impedance reflected into the coil  130  by the coil  124 . At the frequency  168 , the voltage reaches a local minimum, corresponding to the maximum reflected impedance which occurs as the frequency of the measuring oscillator  142  matches the resonant frequency of the sensor  118 . 
     In operation, as depicted in  FIG. 5 , the frequency of the measuring oscillator  142  is controlled by a control signal  170  from the summer  154  which includes a low frequency component from the low frequency source  146 . The control signal  170  thus modulates the capacitance of the variable capacitor  140  resulting in a modulated frequency of the measuring oscillator  142 . By way of example, if the frequency of the measuring oscillator  142  is matched with the resonant frequency of the sensor LC tank, then, as shown in  FIGS. 5 and 6 , the frequency of the measuring oscillator  142  is at a frequency associated with the reference line  172  and the voltage  180  of the measuring oscillator  142  is at the minimum value of the dip  166 . 
     As the control signal  170  increases from a base value at reference line  174  to a maximum value at reference line  176 , the resonant frequency of the measuring oscillator  142  increases from the frequency associated with the reference line  172  to the frequency associated with the reference line  178 . Accordingly the measured voltage of the measuring oscillator  142  increases from an initial value (V 0 ) to a higher value. As the control signal decreases from the maximum value at reference line  176  to the base value at reference line  182 , the resonant frequency of the measuring oscillator  142  decreases from the frequency associated with the reference line  178  to the frequency associated with the reference line  172 . Accordingly the measured voltage of the measuring oscillator  142  decreases from the maximum value to the initial value (V 0 ). 
     The control signal  170  then decreases from the base value at reference line  182  to a minimum value followed by a return to the base value at reference line  184 . Accordingly, the resonant frequency of the measuring oscillator  142  decreases from the frequency indicated by the reference line  172  to the frequency indicated by the reference line  186  and then returns to the frequency associated with the reference line  172 . Accordingly the measured voltage of the measuring oscillator  142  increases from the initial value (V 0 ) to the same maximum value discussed above and then returns to the base value. 
     The line  180  of  FIG. 5  depicts the measured voltage throughout the foregoing sequence. The line  180  thus represents the gain signal that is provided as an input to the demodulator  144  (see  FIG. 3 ). The demodulator  144  further receives a low frequency signal from the low frequency source  146  through the optional phase corrector  148 . The demodulator  144  uses the phase corrected low frequency signal to perform a transfer function on the gain signal (voltage  180 ), as discussed more fully below, and the output of the demodulator  144  is filtered by the filter  150 . 
     As shown in  FIG. 5 , the gain  180  from the reference line  174  to the reference line  182  is identical to the gain  180  from the reference line  182  to the reference line  184 . The phase corrected low frequency signal between the reference line  174  and the reference line  182  (which may be depicted in the same fashion as the control signal  170 ), however, is  180  degrees in phase relative to the phase corrected low frequency signal from the reference line  182  to the reference line  184 . Accordingly, when the phase corrected low frequency signal is used to perform a transfer function on V 0 , the output of the filter  150  is zero. 
     The output of the filter  150  is used to modify the controller signal from the PID controller  152 . Since the output of the filter  150  is “zero” in this scenario, the controller signal out of the PID controller  152  is not changed. Thus, the frequency of the measuring oscillator  142  is maintained at the frequency associated with the reference line  172 . Therefore, when the measuring oscillator  142  is centered on the resonant frequency of the sensor  118 , there is no change in the controller signal generated by the PID controller  152 . 
     If the frequency of the measuring oscillator  142  is lower than the resonant frequency of the sensor  118 , then, as shown for example in  FIGS. 5 and 7 , the frequency of the measuring oscillator  142  is at a frequency associated with the reference line  190  and the voltage  194  of the measuring oscillator  142  is at a value greater than the minimum value associated with the dip  166 . 
     In this situation, as the control signal  170  increases from a base value at reference line  174  to a maximum value at reference line  176 , the resonant frequency of the measuring oscillator  142  increases from the frequency associated with the reference line  190  to the frequency associated with the reference line  192 . Accordingly the measured voltage  194  of the measuring oscillator  142  initially decreases from the initial value (V 0   1 ) to a minimum value associated with the dip  166 . 
     The control signal  170 , however, continues to increase after the measured voltage  194  has decreased to the minimum value associated with the dip  166 . Accordingly, the voltage  194  begins to increase (see  FIG. 7 ). As the control signal  170  decreases from the maximum value at reference line  176  to the base value at reference line  182 , the resonant frequency of the measuring oscillator  142  decreases from the frequency associated with the reference line  192  to the frequency associated with the reference line  190 . Accordingly the voltage  194  decreases to below the V 0   1  and then increases back to the V 0   1 . 
     The control signal  170  then decreases from the base value at reference line  182  to a minimum value followed by a return to the base value at reference line  184 . Accordingly, the resonant frequency of the measuring oscillator  142  decreases from the frequency indicated by the reference line  190  to the frequency indicated by the reference line  196 . The voltage  194  thus increases to a maximum value and then returns to V 0   1 . Throughout this phase, the voltage  194  is greater than the V 0   1 . 
     In the foregoing scenario, the average of the gain  194  from the reference line  174  to the reference line  182  is slightly positive while the average of the gain  194  from the reference line  182  to the reference line  184  is significantly positive as depicted in  FIG. 5 . Thus, the overall signal generated by the transfer function using the phase corrected low frequency signal in the manner discussed above is a large negative signal. Accordingly, the output of the filter  150  is negative. 
     The negative output of the filter  150  is then used to modify the controller signal from the PID controller  152  so as to increase the resonant frequency of the measuring oscillator  142  toward the frequency associated with the local minimum of the dip  166 . Thus, when the measuring oscillator  142  is centered on a frequency below the resonant frequency of the sensor  118 , the controller signal generated by the PID controller  152  controls the measuring oscillator  142  toward a higher frequency. 
     In contrast, if the frequency of the measuring oscillator  142  is higher than the resonant frequency of the sensor  118 , such as is shown in  FIGS. 5 and 8 , the frequency of the measuring oscillator  142  is at a frequency associated with the reference line  200  and the voltage  204  of the measuring oscillator  142  is at a value greater than the minimum value associated with the dip  166 . 
     The measuring oscillator  142  in this scenario, in contrast to that of  FIG. 7 , is operating on the portion of the dip  166  to the right of the dip minimum. Thus, as the control signal  170  increases from a base value at reference line  174  to a maximum value at reference line  176 , the resonant frequency of the measuring oscillator  142  increases from the frequency associated with the reference line  190  to the frequency associated with the reference line  202 . Accordingly the measured voltage  204  of the measuring oscillator  142  constantly increases from the initial value (V 0   2 ) during this phase. As the control signal  170  decreases back to the base value at reference line  182 , the voltage  204  decreases to the V 0   2 . 
     The control signal  170  continues to decrease from the base value at reference line  182  to a minimum value. Accordingly, the resonant frequency of the measuring oscillator  142  decreases from the frequency indicated by the reference line  200  to a minimum value associated with the dip  166 . The control signal  170  continues to decrease after the measured voltage  204  has decreased to the minimum value associated with the dip  166 . Accordingly, the voltage  204  begins to increase until the control signal  170  reaches a minimum value. The control signal  170  then increases back to the base value and the voltage  204  is driven below the V 0   2  and then increases back to the V 0   2 . 
     In this scenario, the average of the gain  204  from the reference line  174  to the reference line  182 , as indicated by  FIG. 5 , is a large positive value while the average of the gain  204  from the reference line  182  to the reference line  184  is slightly positive. Thus, the overall signal generated by the transfer function using the phase corrected low frequency signal in the manner discussed above is a large positive signal. Accordingly, the output of the filter  150  is positive. 
     The positive output of the filter  150  is then used to modify the controller signal from the PID controller  152  so as to decrease the resonant frequency of the measuring oscillator  142 . Thus, when the measuring oscillator  142  is centered on a frequency above the resonant frequency of the sensor  118 , the controller signal generated by the PID controller  152  controls the measuring oscillator  142  toward a lower frequency. 
     Thus, the electronic assembly  120  automatically controls the measuring oscillator  142  to the resonant frequency of the sensor LC tank while the resonant frequency of the sensor LC tank in turn varies with the change of pressure in the sensor compartment  112 . Accordingly, the frequency of the measuring oscillator  142 , which is the same as the resonant frequency of the sensor LC tank, is identified by the frequency counter  156  and available for output to another system or component. 
     The frequency counter  156  may include a precise frequency reference such as a crystal oscillator to facilitate accurate frequency determination. Alternatively, the capacitor value of the variable capacitor  140  may be used to identify the frequency of the measuring oscillator  142 . In the event that a switched capacitor array is used as the variable capacitor  140 , the capacitor value may be available as a digital value. In this embodiment, a reference frequency is not needed. 
     Accordingly, the sensor assembly  100  provides data associated with the pressure within the sensor compartment  112  while protecting the electronic assembly  120  from harsh temperatures or fluids, which may be a liquid or a gas, within the sensor compartment  112 . 
     The sensor  118 , in contrast, is positioned within the sensor compartment  112  and thus exposed to a potentially harsh environment. Protection for the sensor  118  may be provided by a coating such as a passivation layer, a protective gel, etc. In one embodiment, the sensor is made exclusively from materials that resist adverse reactions from aggressive media. Such materials may include silicon, platinum, and gold. 
     In the embodiment of  FIG. 2 , the sensor coil  124  is made of a material exhibiting a change in resistivity as a function of temperature. The use of a material having a resistance that is temperature dependent allows the temperature of the coil  124  to be ascertained. 
     Temperature identification is provided by observing the operation of the electronic assembly  120  when the measuring oscillator  142  is at the resonant frequency of the coil  124  or at least within the dip  166 . In one approach, the change in the measured voltage of the measuring oscillator  142  as the low frequency signal is applied is identified. As discussed above with respect to  FIG. 4 , the dip  166  is caused by the reflection of the impedance of the coil  124  into the coil  130 . The shape of the dip is thus a function of the impedance of the coil  124 , which is inversely related to the Q-factor of the sensor  118 , and the impedance of the coil  124  changes in response to the temperature of the coil  124 . Specifically, as the impedance of the coil  124  increases, the frequency width of the dip  166  increases and the depth of the dip  166  is reduced. 
     Accordingly, for a known change in frequency, such as the change caused by the low frequency signal applied to the measuring oscillator  142 , as the impedance increases, a smaller amplitude of voltage change is observed for the measuring oscillator  142 . Therefore, changes in the amplitude of the voltage excursion provide an indication of the temperature of the sensor  118 . 
     Alternatively, the voltage of the measuring oscillator  142  may be used. Specifically, since the minimum value of the dip  166 , which is the filtered signal out of the filter  150 , is related to the impedance of the coil  124 , changes in the minimum voltage observed provide an indication of the temperature of the sensor  118 . 
     Other mechanisms may influence the accuracy of the sensor assembly  100  in addition to temperature. One such mechanism is present when the measured voltage of the sensor assembly  100  is frequency dependent. For example, as seen in  FIG. 4 , the voltage peaks at about reference lines  162  and  164  and falls off not only in the dip  166 , but also at frequencies higher than the frequency associated with the reference line  164  as well as at frequencies below the frequency associated with the reference line  162 . Therefore, if the measuring oscillator  142  is not operating in the frequencies associated with the dip  166 , the measuring oscillator  142  may not track into the dip  166 . Accordingly, an algorithm or other mechanism is preferably provided to ensure that the sensor assembly  100  is operating in the dip  166 , and not at another location along the line  160 . 
     Additionally, when the sensor assembly  100  applies the low frequency signal to the variable capacitor  140 , the resultant voltage or gain is ultimately transmitted to the PID controller  152  resulting in a modified control signal from the summer  154 . The low frequency signal thus adds some amount of noise to the sensor assembly  100 . The system  210  depicted in  FIG. 9  mitigates this type of noise. 
     The system  210  includes a measuring oscillator  212  and a control system  214 . The measuring oscillator  212  includes a primary coil  216  and a variable capacitor  218 . The system  210  further includes a capacitor  220 , a gain variable transconductance amplifier (gvOTA)  222 , and an automatic gain control  224 . The gvOTA  222  functions as a feedback voltage controlled current source. The feedback voltage control is provided by the capacitive voltage divider of the variable capacitor  220  and the capacitor  222 . 
     In operation, the control system  214  maintains the amplitude of the system  210  constant by controlling the gain on the gvOTA  222 . Thus, if the control system  214  senses a decrease in the frequency, a signal is provided to the variable capacitor  218  reducing the capacitance of the variable capacitor  218 . Accordingly, the resonant frequency of the measuring oscillator  212  increases and less current is needed to drive the measuring oscillator  212 . At the same time the frequency is modified, the voltage between the variable capacitor  218  and the capacitor  220  is reduced since the capacitance of the variable capacitor  218  has been reduced. 
     The reduced voltage is felt by the gvOTA  222  causing less current to be provided to the AGC  224 . Thus, instead of forcing the AGC  224  to react and force the current output of the gvOTA  222  to be reduced, modification of the capacitance of the variable capacitor  218  functions to change the resonant frequency of the measuring oscillator  212  and to also reduce the current output of the gvOTA  222 . Accordingly, noise associated with introduction of a low frequency signal is reduced. 
     Components selected for the sensor  100  may implicate other mechanisms influencing the accuracy of the sensor  100 . For example, the variable capacitors  140 / 220  may be varactor diodes. A digital-analog converter (DAC) could be incorporated in varactor diode system to allow use of a digital control system. In another embodiment, the variable capacitors  140 / 220  employ a switched capacitor technology. A switched capacitor system may incorporate binary weighted capacitors (e.g., C-2C arrangements). Unity caps may be further incorporated to minimize matching problems. 
     A binary weighted C-2C system may, however, generate undesired charging and discharging of capacitances. For example, as the capacitor array approaches and passes the ½ full range value, the smaller capacitors which are at full range value as the ½ value is approached are discharged and the largest capacitor, which is not charged when the system is below ½ full range value, is charged. Thus, the amplitude control (AGC) is subjected to an injection of noise which affects the sensor signal in general. 
     One approach to reduce capacitor related noise is the incorporation of a switched capacitor such as the switched capacitor  230  as depicted in  FIG. 10 . The switched capacitor  230  includes a decoder  232 , a plurality of switching transistors  234 , and a respective plurality of unity capacitors  236 . Each of the switching transistors  234  is individually controlled through the decoder  232 . The switched capacitor  230  can thus be directly controlled from a digitally implemented control. 
     The switching transistors  234  may be N-MOS transistors. By way of example, the switching transistors  234  may be positioned within a well (not shown). The well may be biased with a voltage to make optimum use of the switching transistors by reducing resistance. The incorporation of level shifters for controlling the gate voltage of the switch transistors reduces the resistance of the system. 
     In a switched capacitor system, the voltage across the non-variable capacitor (e.g., the capacitor  220 ) must be smaller than the threshold voltage of the switch transistors of the variable capacitor. A system that provides a good voltage control for the non-variable capacitor  280  is depicted in  FIG. 11 . The measuring oscillator  250  in  FIG. 11  includes a gvOTA  252 , an OPA  254 , and a PID OPA  256 . One input  258  of the gvOTA  252  is connected to a constant voltage source. One embodiment of such a voltage source is shown in  FIG. 11  as the outlet  260  of an OPA  262 . One input  264  of the OPA  262  is biased by a voltage divider provided by a resistor  266  and a resistor  268 . The other input  270  of the OPA  262  is connected to the output  260  of the OPA  262 , through a capacitor  272 , forming a voltage follower. The capacitor  272  is optional, and typically used for a selected OPA  262  that is not as strong or fast as desired for a particular circuit. 
     The second input  274  of the gvOTA  252  is biased by a capacitive divider provided by a variable capacitor  276 . The variable capacitor  276  includes a switch  278 , which may be one of a plurality of switched capacitors within a well in a VDD 2 , and a capacitor  280 . A resistor  282  is in parallel with the capacitor  280  between the input  258  and the input  274  of the gvOTA  252 . The variable capacitor  276  and the capacitor  280  are in parallel with a coil  284 . The coil  284  functions as the primary coil for coupling with a sensor coil such as the coil  124 . One side of the coil  284  is connected to an output  286  of the gvOTA  252  and the other side of the measuring oscillator is connected to an output  288  of the gvOTA  252 . 
     The output  288  of the gvOTA  252  is further connected to an N-transistor  290  and to a P-transistor  292 . The N-transistor  290  is configured to charge a capacitor  296  while a weak current source  298  is positioned in parallel with the capacitor  296  to discharge the capacitor  296 . The P-transistor  292  is connected to an input  300  of the OPA  254  and is further configured to discharge a capacitor  302 . A weak current source  304  is positioned in parallel with the capacitor  302  to charge the capacitor  302 . 
     The output  306  of the OPA  254  in this embodiment is connected to a second input  308  of the OPA  254  and, through a resistor  310 , to the second input  312  of the PID OPA  256 . The output  314  of the PID OPA  256  is connected through a capacitor  316  and a resistor  318  to the second input  312  of the PID OPA  256 . The output  314  of the PID OPA  256  also controls the gain of the gvOTA  252 . 
     The transistors  290 / 292 , capacitors  296 / 302  and current sources  298 / 304  are configured to apply the highest voltage output of the gvOTA  252  to the input  294  of the PID OPA  256  and to apply the lowest voltage output of the gvOTA  252  to the input  300  of the OPA  254 . This is accomplished because the transistor  290  rapidly charges the capacitor  296  while the current source  298  slowly discharges the capacitor  296 . Accordingly, the capacitor  296 , which is connected to the input  294 , is kept at the highest output voltage of the gvOTA  252  minus the threshold voltage of the transistor  290 . 
     Additionally, the transistor  292  rapidly discharges the capacitor  302  while the current source  304  slowly charges the capacitor  304 . Accordingly, the capacitor  304 , which is connected to the input  300 , is kept at the lowest output voltage of the gvOTA  252  plus the threshold voltage of the transistor  292 . Alternatively, sample and hold circuitry may be provided to apply the highest voltage output of the gvOTA  252  to the input  294  of the PID OPA  256  and to apply the lowest voltage output of the gvOTA  252  to the input  300  of the OPA  254 . 
     Accordingly, when operating with the voltage of the capacitors  296  and  302  matched, the peak to peak voltage of the measuring oscillator comprising the coil  284  and the capacitors  276  and  280  equals the average threshold values of the transistors  290 / 292 . Thus, the threshold values of the transistors  290 / 292  can be selected to establish the desired voltage applied across the switch  278 . The DC portion of the voltage between the capacitor  276  and the capacitor  280  is further established by the resistor  282 . In one embodiment, the resistor  282  is a high ohmic resistor. Alternatively, a weak current source may be used to establish a desired voltage between the capacitor  276  and the capacitor  280 . 
     A number of modifications may be incorporated with the foregoing circuits in addition to those identified above. By way of example, the PID controller circuit around OPA  256  may be replaced by a PI controller circuit. Additionally, an adjustable controller may be used so as to optimize the sensor system  250  for a particular sensor coil or frequency range. The gvOTA  252  may also be provided with a gain range switching capability to allow optimization of the sensor system  250  for use with different sensor elements or temperature regimes. Additionally, other components may be incorporated into the sensor system  250  for a particular application. For example, a linearizer circuit may be incorporated to linearize the voltage gain of the gvOTA  252 . 
     In an ideal scenario, the inclusion of the gvOTA  252  provides a frequency independent response. Typically, however, a gvOTA will exhibit some frequency dependent characteristics. For example, the gain of a gvOTA typically rolls off at higher frequencies. Accordingly, more gain is needed for a measuring oscillator at higher frequencies. The sensor system  350  of  FIG. 12  mitigates this effect. 
     The sensor system  350  includes many of the same components as the system  250  which are numbered in like manner as the components of the system  250 . One difference between the system  250  and the system  350  is that in place of the gvOTA  252 , the system  350  includes a gvOTA  352 . The gvOTA  352  includes inputs  354  and  356  which are configured similarly to the inputs  258  and  274  of the system  250 . The output  358  of the gvOTA  352 , however, is not connected to the input  354 . Additionally, the gvOTA  352  includes two additional inputs, input  360  and  362 . The input  360  is connected to the output  358  while the input  362  and the input  354  are connected to the lower leg of the LC tank with the voltage at this leg established by the output  260  of the OPA  262 . 
     The configuration of the gvOTA  352  is thus modified to allow differential voltages between inputs to be summed inside of the gvOTA  352  with a fixed or variable ratio as desired. In one embodiment, the input of the gvOTA  352  is configured as a standard differential N-MOS input stage employing two input transistor pairs in parallel with a defined transistor ratio as depicted in  FIG. 13 . The input  370  of  FIG. 13  includes input stage transistors  372 ,  374 ,  376 , and  378 . 
     The input transistors  372  and  378  are configured as one input pair and the input transistors  374  and  376  are configured as a second input pair. Signals generated by the paired transistors  372 ,  374 ,  376 , and  378  are summed at the drain connection of the differential amplifier stage. The ratio of the summation is defined by the W/L ratio of the input transistors relative to each other. By way of example, if (at the same length) the width of the input of transistor  374  is twice the width of the input of transistor  372 , then a signal at the input of transistor  374  will get amplified as twice as much as an input signal on the input of transistor  372 . Therefore, the gain of the gvOTA  252  can be established by selecting the ratio of C 1 /C 2  and the W/L of the input of transistor  372  and “input of transistor  374 . 
     The gain of the capacitive divider (the variable capacitor  276 ) can also be adjusted by the capacitor  400 , which in the system  350  is a trimable capacitor. The effect of changing the capacitance of the capacitor  350  is depicted in  FIG. 14 .  FIG. 14  depicts a graph of the gain of the gvOTA  352  verses frequency without any coupling between the coil  284  and a sensor coil. The measuring oscillator frequency response curves  404 ,  406 ,  408 ,  410 , and  412  reflect the effects of changing the capacitance from a lower value (curve  404 ) to iteratively higher values, with curve  412  representing the highest value. 
     The capacitor  400  can thus modify the measuring oscillator frequency response from a negative slope (curve  404 ), to a substantially flat slope (curve  410 ), to a positive slope (curve  412 ). By selecting the capacitance associated with the slope  410  and coupling the coil  284  with the coil  124 , the measuring oscillator frequency response curve  416  of  FIG. 15  is obtained. The desired slope can thus be selected such so as to obtain the desired operating characteristics when the coil  284  is coupled with a sensor coil such as coil  124 . 
     While the present invention has been illustrated by the description of exemplary system components, and while the various components have been described in considerable detail, applicant does not intend to restrict or in any limit the scope of the appended claims to such detail. Additional advantages and modifications will also readily appear to those skilled in the art. The invention in its broadest aspects is therefore not limited to the specific details, implementations, or illustrative examples shown and described. Accordingly, departures may be made from such details without departing from the spirit or scope of applicant&#39;s general inventive concept.