Abstract:
A low frequency amplifier uses a switched bridge circuit, providing a first frequency output. A transformer circuit receiving the first frequency output from the switched bridge circuit. Power from the transformer is output from a plurality of secondaries and the power from the secondaries is supplied to the corresponding output switching circuits and provided as switched outputs from the transformer circuit. The switched outputs from the transformer circuit are responsive to a transformer output from the transformer at the first frequency, and switch the transformer outputs in a timed sequence to provide a combined second frequency output. The second frequency output has a lower frequency than the transformer outputs.

Description:
FEDERALLY-SPONSORED RESEARCH AND DEVELOPMENT 
   This invention was developed with funds from the United States Department of the Navy. Licensing inquiries may be directed to Office of Research and Technical Applications, Space and Naval Warfare Systems Center, San Diego, Code 2112, San Diego, Calif., 92152; telephone 619-553-2778; email: T2@spawar.navy.mil. 

   BACKGROUND 
   1. Field of the Invention 
   The disclosed subject matter relates to an amplifier design, of a type using magnetic energy. The disclosed subject matter also relates to low frequency power amplifiers. 
   2. Background of the Invention 
   In design of low frequency power amplifiers, it is desired to provide low frequency power amplification in a manner that minimizes amplifier size and weight. Current low frequency power amplifiers are relatively large and heavy. This is because present amplifiers contain magnetic components which must provide operational function at the output signal frequency. Examples of magnetic devices used in the magnetics are transformers and inductors. 
   The size of a magnetic device (transformer or inductor) varies to a first approximation linearly and inversely with frequency, which means that a magnetic device operating at a higher frequency may be manufactured to be much smaller than one having comparable performance characteristics, but operating at a lower frequency. Thus for a low frequency power amplifier, the size and weight of the magnetics will be many times greater than if the amplifier could be built with all magnetics operating at a switching frequency which was orders of magnitude higher than the desired output signal frequency. 
   U.S. Pat. No. 5,815,384, to Hammond, et al. describes a transformer circuit in which an AC switching stage preferably generates pulses at a frequency which may be up to several orders of magnitude higher than a line frequency, typically 60 or 400 Hz and having a duty cycle which may be up to about 98-99% of the pulse period. The circuit includes a transformer which requires less core volume and mass than conventional transformers, and includes a rectifier for transforming a first time varying input signal, such as a sinusoid or saw tooth signal, into a full-wave rectified voltage signal. 
   SUMMARY 
   A low frequency amplifier is configured to use magnetic elements at a frequency which may be higher than that of the amplified output frequency of the amplifier. A switched bridge circuit provides a first frequency output. A transformer circuit receives the first frequency output from the switched bridge circuit, which results in the transformer operating at the frequency of the first frequency output. A switched output from the transformer circuit is provided and is responsive to a transformer output from the transformer at said first frequency. The transformer output is switched to provide a first sense output and a second sense output in a timed sequence and to provide a second frequency output. In the case of the transformer operating at a higher frequency, the second frequency output has a lower frequency than the first frequency output. 

   
     BRIEF DESCRIPTION OF THE DRAWING 
       FIG. 1  is a schematic diagram showing an example amplifier. 
       FIG. 2  is a depiction of a pulse train, corresponding to pulses across the output of the bridge. 
       FIG. 3  is viewed during the beginning of a series of positive pulses, but the opposite polarity effect occurs under negative pulses. 
       FIGS. 4A and 4B  are diagrams showing an implementation of back-to-back FETs. 
       FIG. 5  is a diagram showing an equivalent circuit of the low power amplifier of  FIG. 1 . 
       FIG. 6  is a schematic block diagram showing a control circuit implemented with a lookup table (LUT). 
       FIG. 7  is a schematic diagram showing an example amplifier, in which a series output arrangement is used to reduce the voltage parameters of switching components. 
   

   DETAILED DESCRIPTION 
   A low frequency power amplifier is configured to employ only higher frequency magnetics. This reduces the size of the magnetics, and thereby minimizes amplifier size and weight. The output of the magnetics is provided to a down-convertor, which switches HF signal inputs to simulate a LF output. Pulses in the down-convertor are selectively rectified by switches to steer positive or negative pulses to desired outputs. 
     FIG. 1  is a schematic diagram showing an example amplifier. Energy is supplied by DC source  111 . Additional optional energy storage is provided by energy storage capacitor  123 , which may be by way of example, an ultra capacitor or a conventional capacitor. A full bridge  128  with switches  131 - 134  is operated to provide a pulse width modulated (PWM) pulse train for the output signal across amplifier primary  142  and capacitor  141 , forming an LC circuit. The output across LC circuit  141 ,  142  is applied across the transformer (through primary  141 ) and is the output that is to be amplified. 
   The DC source  111  can be any convenient power source, such as a battery, outside power supply or the equivalent. With appropriate rectification or switching, an AC power supply can also be used. 
   Also shown are two secondary coils  147 ,  148 , each of which is connected to switching rectifier inverter output circuits  151 ,  152 . Inverter output circuit  151  comprises switches  161 - 164  and inverter output circuit  152  comprises switches  165 - 168 . Each of switches  161 - 168  is series connected to diodes  171 - 178 , respectively. A control circuit  180  is used to control switches  131 - 134  and switches  161 - 168 . The control of the switches  131 - 134  and of switches  161 - 168  is independent, meaning that the switches are typically not turned “on” or “off” simultaneously. 
   The use of two secondary coils  147 ,  148  and separate inverter output circuits  151 ,  152  allows a series voltage output between inverters  151 ,  152 , so that switches  161 - 168  and diodes  171 - 178  can have circuit parameters which are approximately half that required for if there were a single inverter output circuit providing the entire output. This is particularly advantageous because of the maximum voltage parameters for commonly available field effect transistor devices used in inverters  151 ,  152 . The use of two inverters  151  is given as an example of multiple inverters, and it is further possible to provide a greater number of secondary coils and inverter output circuits, in order to provide a proportional increase in the output voltage. 
   A further description of the inverters  151 ,  152  is found in U.S. Pat. No. 5,815,384, to Hammond, et al., and which is incorporated by reference herein. One advantage of the use of switching techniques, the amplifier may be operated to provide a controlled output voltage signal having an adjustable amplitude that is not completely determined by the turns ratio between the primary and secondary windings of the transformer. In particular, inverters  151 ,  152  are connected to provide a series output, through LC circuit elements  181 - 184 , at nodes  187 ,  188 . 
   The outputs of the inverters  151 ,  152  are capacitively coupled using LC circuits  181 ,  182 , and  183 ,  184 , to provide the output signal at nodes  187 ,  188 . An LC output load  191 ,  192  provide an output balance. Also shown is an equivalent L-C-R output circuit  196 - 198 . 
     FIGS. 2 and 3  are depictions of pulse trains.  FIG. 2  shows the pulse train, corresponding to pulses across the output of the bridge  128 . Referring to  FIG. 1  with reference to  FIG. 2 , the pulses from bridge  128  are formed positive when switches  131 + 134  are “on” (closed) and negative when switches  132 + 133  are “on”. These pulses are then selectively rectified by turning switches  161  and  163  “on” (closing  161  and  163 ) to steer the first positive pulse to output node  187  and the turning  162  and  164  “on” to steer the first negative pulse to output node  188 . Using this repeating switch scheme, the voltage across nodes  187 ,  188  takes the form of a series of pulses  201 - 206 . Pulses  201 - 206  are switched by output switches  161 - 168  to provide rectified half wave output pulses  301 - 306 , depicted in  FIG. 3 . The output pulses  301 - 306  are smoothed to a sine wave, appearing in  FIG. 3  as a portion of a sine wave  311 . By operation of switches  161 - 168 , the output is inverted, to generate a full sine wave.  FIG. 3  is viewed during the beginning of a series of positive pulses, but the opposite polarity effect occurs under negative pulses. Referring again to  FIG. 1 , the output filter LC filters  181 ,  182  and  183 ,  184  filter this waveform and averages or extracts the fundamental component to form voltage, at output nodes  187 ,  188 . 
   Switches  161  and  162  are operated in unison and operate at close to a 50% duty cycle. The reason for this will become clear when the reactive energy concerns are discussed. Although, for rectification of the first positive pulse, only  161  is conducting the pulse&#39;s energy. Similarly to rectify the first negative pulse  161  and  162  are “on” at close to a 50% duty ratio (on almost half of the output frequency period). 
   A similar situation exists for the negative half cycle of the signal to be amplified. In the negative half-cycle, the switches  161  and  163  are going to be turned “on” for positive pulses coming from the transformer. Likewise switches  162  and  164  are going to be turned “on” to steer a negative transformer pulse to help form the negative half cycle of the output waveform. 
   Control circuit  180  is used to control switches  131 - 134  and switches  161 - 168  in order to provide the desired output frequencies from bridge  128  and from inverters  151 ,  152 . An example of such a control circuit is described in the aforementioned U.S. Pat. No. 5,815,384, to Hammond, et al. In operation, closing two switches of opposite polarity, meaning  131 ,  134  or  132 ,  133 , results in current flow through primary coil  141 . The operation of switches  131 - 134  thereby controls the pulse width of pulses  201 - 206  ( FIG. 2 ). 
     FIGS. 4A and 4B  are diagrams showing an implementation of back-to-back FETs  401 ,  402 ,  411 ,  412 . Switches  161 - 164  and  165 - 168  with their diodes  171 - 174  and  175 - 178  could also be implemented as the back-to-back FETs  401 ,  402 ,  411 ,  412  with higher quality bypass diodes  421 ,  422 ,  431 ,  432 . This establishes parallel bridge and rectifier circuits  441 ,  442 ,  445 ,  446 , which can be used to further step the output voltage. 
   Referring back to  FIG. 1 , switches  161 - 168  are configured to operate in pairs, the pairs being switches  161  and  162 , switches  163  and  164 , switches  165  and  166 , and switches  167  and  168 . This is done for convenience of circuit design, since diodes  171 - 178  function to direct the current flow through switches  161 - 168 . Referring back to  FIGS. 4A and 4B , the particular design of FETs is such that the switching function for the separate switches in the pairs of switches  161 - 168  is easily separated. This also halves the current flow through the individual switches  161 - 168 . By way of non-limiting example, the switches of each pair are opened and closed simultaneously with the other switch in the pair. In other words, simultaneous operation is effected for switches  161  and  162 , for switches  163  and  164 , for switches  165  and  166 , and for switches  167  and  168 . 
   The arrangements of back-to-back FETs  401 ,  402 ,  411 ,  412  are useful for low frequency applications, such as driving sonar transducers. Here voltages of 2500 volts are frequently necessary. Referring to  FIG. 1 , although a single transformer T 1  could step the voltage up to this level, this may exceed the rating of switches  141 ,  142 ,  151 ,  152  that exists for commonly available FETs. The rating of the FETs is a design concern because FETs are the preferred switching element due to ease of drive and speed of transition. It is difficult to find FETs with voltage ratings much past 1200 v and sonar applications often require 2500 v. Thus it would be possible to operate 3 bridge rectifier circuits each with an output voltage of around 800 v or higher, and then to capacitively sum their voltages for the required 2500 volts. If insulated-gate bipolar transistors (IGBTs) are used to implement switches  161 - 168 , their higher voltage rating (˜2500 v) would allow fewer stages to be used, for example 2 stages. 
     FIG. 5  is a diagram showing an equivalent circuit of the low power amplifier of  FIG. 1 . A problem to be considered with this approach to amplification is reactive energy. Sonar transducers are highly capacitive and a path back to the DC source  111  must be provided. Essentially the function of this amplifier circuit is to provide a source that looks like that depicted in  FIG. 5 . A signal generator  511  provides an output through LC circuit  513 ,  514 , which functions as an LC averaging filter. The output is then received by transducer  521 , which in a non-limiting example could be a sonar transducer. The sonar transducer  521  is given as a typical reactive load, but the load  521  but may be any suitable load. The reactive load given as an example is considered reactive in that current and voltage is drawn out of phase. The output load has an equivalent circuit which is represented by capacitor  531  in parallel with inductor  541 , capacitor  542  and resistor  543 . 
   During the “on” period of the bridge switches ( 131   134 ,  FIG. 1 ), a path exists from the load through one of the “on” rectifier switches through the transformer, through the “on” bridge switches  131 - 134  and into the DC source  111 . During the “off” times for the bridge switches  131 - 134 , the path for reactive current is an open circuit unless some other action is taken. For the amplifier to simulate the above ideal source, the bridge switches  131 - 134  would have to provide a short or zero voltage state for the reactive current. This can be done in the following way. 
   If, for example, switches  131 ,  134  are “on”, thus delivering a positive pulse, at the completion of the pulse, switch  131  is turned “off”. After switches  131  is “off”, switch  132  is turned “on”. This then allows a shorted path for current to flow. When it is time for a negative pulse to be produced by the bridge, switch  134  is turned “off”. When switch  134  is “off”, switch  133  is turned “on” and a negative pulse is produced. A similar routine then occurs at the end of the negative pulse to provide a shorted path for reactive current. 
   An example of a control circuit  180  is depicted in  FIG. 6 , and comprises a clock  601 , counter  608 , and lookup table (LUT)  611 . In the example, output counter  608  is an 8 output counter, and LUT  611  is implemented as a 256×8 ROM. 
   Control of switches  131 - 134  and switches  161 - 168  is accomplished in response to control circuit  180 : An output of clock  601  is inputted into counter  608  which then addresses LUT  611 . As this clock counter system goes through all 256 addresses, one switching cycle is executed. 
   By way of example, given a 25 kHz period, the switching cycle would be approximately 40 ms. For the  161 / 162  and  171 / 172  switches the description is straight forward.  161 / 162  are “on” for the positive half cycle (128 clock pulses) and  171 / 172  “on” for the negative half cycle (128 clock pulses). Actually there is a short dead time (−100 ns) between turning  161 / 162  “off” and  171 / 172  “on” and vice versa. 
   A more interesting aspect is determining what the states of the bridge switches  131 - 134  should be. For a desired 1 kHz switching frequency there would be 130 switch periods in an output signal waveform. An even number of switch periods might be preferred as this would tend to drive the bridge transformer equally in the positive and negative direction, although there is a capacitor in series with the bridge transformer primary to prevent transformer saturation. There would potentially be a set of 30 ROM LUTs to be sequentially stepped through to complete a full PWM cycle at 1 kHz. Because the positive and negative half-cycles are symmetric 15 ROM LUTs could probably be used. 
   Still, because this is a relatively large number, it might be viewed as stepping through one LUT whose individual values are a function of time. Each LUT value would cycle through 15 potential changes in value before repeating. Such an approach could advantageously be implemented in software. 
     FIG. 7  is a schematic diagram showing an example amplifier, in which a series output arrangement is used to reduce the voltage parameters of switching components. As is the case with the example of  FIG. 1 , the front end consists of DC source  111  and energy storage capacitor  123 , providing power full bridge  128  with switches  131 - 134 . The output is provided to LC circuit  141 ,  142 . 
   A plurality of transformer secondaries  721 ,  722 , . . .  729  provide outputs to switching rectifier inverter output circuits  751 ,  752 , . . .  759 , which are arranged in series. Switching inverter circuits  751 ,  752 , . . .  759  are isolated by capacitors  771 , and by inductors  781 . The result is that each inverter circuit  751 ,  752 , . . .  759  is able to include switching and rectifier components (e.g.,  161 - 168  and diodes  171 - 178  in  FIG. 1 ) having voltage ratings which are a fractional proportion to the output voltage between output nodes  187 ,  188 . Inductors  781  also reduce transients in the switching outputs across the inverter circuits  751 ,  752 , . . .  759 . 
   By way of non-limiting example, if nine inverter circuits  751 ,  752 , . . .  759  are used, and total peak-peak voltage across output nodes  187 ,  188  is 3500 volts, then the peak-peak voltage across each inverter circuit would be 388 volts. Switching components are more easily available at a rated voltage of 500 volts than at 3500 volts, so that the ability to use lower voltage components is advantageous. If a 3500 volt peak-peak circuit has an approximate RMS voltage of 2500 volts, the RMS voltage across the nine inverter circuits  751 ,  752 , . . .  759  would be 280 volts. 
   It will be understood that many additional changes in the details, materials, steps and arrangement of parts, which have been herein described and illustrated to explain the nature of the invention, may be made by those skilled in the art within the principle and scope of the invention as expressed in the appended claims.