Abstract:
An automatic ATD control circuit operates with a first delay circuit accepting a system clock pulse as an input and producing a delayed version of the system clock pulse as an output. The delay to the system clock is performed to allow a frequency comparison in a later part of the circuit. An edge detection circuit operates when the delayed system clock is received and senses an edge of the delayed system clock pulse. A pulse output from the edge detection circuit feeds into a second delay circuit; the second delay circuit produces an output pulse where a period of the pulse is determined by delay characteristics of the sense amplifier and is thus independent of system clock frequency. The pulse is compared to the system clock frequency. If the system clock frequency is above a determined frequency, the automatic ATD control circuit is disabled.

Description:
TECHNICAL FIELD 
   The present invention relates generally to operation of nonvolatile memory arrays and specifically to operations of nonvolatile memory arrays over a wide range of operating frequencies with low power consumption below a critical frequency. 
   BACKGROUND ART 
   Non-volatile memory devices, such as electrically erasable and programmable read only memories (EEPROMs), comprise core arrays of memory cells including a variable threshold transistor. Each memory cell can include a number of transistors; at least one of which will be a variable threshold (i.e., programmable) transistor. 
   With reference to  FIG. 1 , a portion  100  of a prior art memory array includes a plurality of memory cells  101 ; each of the plurality of memory cells  101  includes a pair of transistors, a select transistor  101 A and a variable threshold transistor (i.e., a floating gate transistor)  101 B. According to one version of the prior art, the select transistor  101 A is an n-channel enhancement transistor, and the floating gate transistor  101 B is an n-channel native transistor. Other kinds of the plurality of memory cells  101  each including a greater number of transistors are known in the prior art as well. Additionally, various arrangements of the plurality of memory cells  101  are known, such as NAND EEPROM and NOR EEPROM arrays. 
   The plurality of memory cells  101  is each interconnected by a plurality of wordlines lines  103 , a plurality of sense lines  105 , and a plurality of bitlines  107 . In particular, drains of the each of the select transistors  101 A are connected to one of the plurality of bitlines  107 . A gate of each of the select transistors  101 A and the floating gate transistors  101 B is each connected to one of the plurality of wordlines  103  and sense lines  107  respectively. 
   In  FIG. 2 , a non-volatile memory arrangement  200  of the prior art includes a read select transistor  201 , a read select line  201 A, a sense amplifier  203 , a data bus  203 A, and a wordline decoder  205 . The non-volatile memory arrangement further includes one each of the select transistors  101 A and the floating gate transistors  101 B from  FIG. 1 . As was the case in  FIG. 1 , according to an n-channel implementation of the select  101 A and the floating gate  101 B transistors, the drain of the select transistor  101 A will be connected to one of the plurality of bitlines  107 , and respective gates of the select  101 A and the floating gate  101 B transistors are connected respectively to one of the plurality of wordlines  103  and sense lines  105 . The wordline  103  is driven by a word line decoder  205 . 
   The read select transistor  201  is connected to the read select line  201 A. When a read operation is active, the read select transistor  201  is turned on, thereby electrically connecting the bitline  107  to the data bus  203 A. The data bus  203 A, in turn, is connected to the sense amplifier  203 . When the non-volatile memory arrangement  200  is subject to a read operation, a conductive state of the memory cell  101  is queried by connecting the bitline  107  to the sense amplifier  203  and applying appropriate bias voltages to the selected bitline  107 , sense line  105 , and wordline  103 . If the select transistor  101 A is turned on and the bias voltage applied to the sense line  105  exceeds a threshold of the floating gate transistor  101 B, current will flow from the bitline  107  to ground through the memory cell  101  and the sense amplifier  203  will detect a “low” state. Conversely, if the bias voltage applied to the sense line  105  does not exceed the threshold of the floating gate transistor  101 B, then no current will flow through the memory cell  101 , and the sense amplifier  203  will detect a “high” state. While the sensing approach just described provides an operable memory arrangement, power consumption levels which characterize this approach are disadvantageous. 
   Power requirements of a contemporary memory sense amplifier are indicated in the dynamic power requirement, P dyn , as a function of operating frequency, f op , graph  300  of  FIG. 3 . A constant sense amplifier consumed power trace  301  is indicative of a minimum power requirement, per wordline, any time the sense amplifier  203  ( FIG. 2 ) is in an operational mode. A minimum sense amplifier power, P min , is determined by
 
 P   min   =V   dd   ·I   SA  
 
where V dd  is the system voltage and I SA  is the sense amplifier current. A linear expression of total memory array power without sense amplifiers, P array ,  303  is governed by
 
 P   array   =C   core   ·V   dd   2   ·f   op  
 
where C core  is determined from a total gate-source capacitance, C gs , value of each of the memory transistors within the plurality of memory cells  101  ( FIG. 1 ). A total dynamic power requirement  305  is then determined by
 
 P   dyn =( C   core   ·V   dd   2   ·f   op )+( V   dd   ·I   SA )
 
which is merely a summation of the constant amplifier consumed power  301  and the linear expression of total memory array power without sense amplifiers  303 .
 
   The dynamic power, P dyn , is a function of one variable—operating frequency, f op . Other functional dependencies, C core , V dd , and I SA , are all fixed for a given memory array configuration. Therefore, it is desirable to minimize the total dynamic power requirement, especially in situations where either the operating frequency is variable during memory array operation or a given memory array is adaptable to a range of operating frequencies within a given circuit. 
   SUMMARY 
   An automatic address transition detection (ATD) circuit and method is described herein which allows a memory device to operate over a wide range of frequencies; the circuit and method provide for operation under reduced power consumption of the device if the device is operating in accordance with a low system clock frequency (less than, for example, 1 MHz). The reduction in power consumption derives from operating sense amplifiers within the memory device with a steady-state bias current only as compared with a higher-level of current needed for reading a state of memory cells. Therefore, in a system operating at a relatively low clock frequency, the higher-level of current is supplied to the sense amplifiers only at times when they are needed for reading memory cells. 
   The automatic ATD circuit operates, in one embodiment, with a first delay circuit configured to accept a system clock pulse as an input and produce a delayed version of the system clock pulse as an output. The delay to the system clock is performed to allow a frequency comparison in a later part of the circuit. A rising-edge detection circuit operates when the delayed system clock is received and senses a rising-edge of the delayed system clock pulse. A pulse output from the rising-edge detection circuit feeds into a second delay circuit; the second delay circuit produces an output pulse where a period of the pulse is determined by delay characteristics of the sense amplifier and is thus independent of system clock frequency. The pulse is compared to the system clock frequency. If the system clock frequency is above a determined frequency, the automatic ATD circuit is disabled. If the clock frequency is below the determined frequency, the automatic ATD circuit is enabled and provides a sense amplifier enable signal only when a memory cell read is required. 
   In an exemplary embodiment of a method of operating the automatic ATD circuit, steps involve delaying an input system clock signal by a first delay period; generating a first pulse (e.g., a rising-edge pulse) based on a the delayed input system clock signal; determining a second delay period based on delay characteristics of the sense amplifier (i.e., sense amplifier characteristics of a time to turn on, a charge delay time, and a time to turn off); producing a critical signal pulse based on the generated pulse and the determined second delay period; and comparing a first period of the system clock signal to the second delay period of the critical signal pulse. If a result of the comparison determines that the first period is shorter than the second period, an address transition detection (ATD) disable pulse is produced. If a result of the comparison determines that the first period is longer than the second period, an address transition detection (ATD) enable pulse is produced. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a portion of a prior art non-volatile semiconductor memory core arrangement. 
       FIG. 2  is a prior art non-volatile semiconductor memory arrangement, which includes a conventional sense amplifier arrangement. 
       FIG. 3  is a graph indicating dynamic power requirement of a prior art conventional sense amplifier arrangement as a function of operating frequency. 
       FIG. 4  is a graph indicating dynamic power requirements of a sense amplifier arrangement in accordance with the present invention, also indicated as a function of operating frequency. 
       FIG. 5A  is a block diagram of an exemplary embodiment of an automatic address transition detection (ATD) circuit in accord with the present invention as used in a memory circuit. 
       FIG. 5B  is a block diagram of an exemplary embodiment of the automatic ATD circuit of  FIG. 5A . 
       FIG. 5C  is a schematic diagram of an exemplary embodiment of a pulse-edge detector circuit as employed in the automatic ATD circuit of  FIG. 5B . 
       FIG. 5D  is a schematic diagram of an exemplary embodiment of a sense amplifier critical pulse generator circuit as employed in the automatic ATD circuit of  FIG. 5B . 
       FIG. 5E  is a schematic diagram of an exemplary embodiment of a transimpedance amplifier circuit as employed in the sense amplifier of  FIG. 5A . 
       FIG. 6  is an exemplary timing diagram of the automatic ATD circuit of  FIG. 5A . 
       FIG. 7  is an exemplary timing diagram of the automatic ATD circuit of  FIG. 5A  wherein a variable system clock frequency is employed. 
   

   DETAILED DESCRIPTION 
   With reference to  FIG. 4 , power requirements of a memory sense amplifier in accordance with the present invention are indicated in the dynamic power requirement, P dyn , as a function of operating frequency, f op , graph  400 . A constant sense amplifier consumed power trace  401  is indicative of a minimum power requirement any time a sense amplifier is in operational mode, i.e., constantly activated. In the present invention, the sense amplifiers are frequently operating in a residual-power mode, described infra. A residual-power mode, P res , trace  403  is determined by
 
 P   res   =m·V   dd   ·I   SA,bias  
 
where m is an integer related to a total number of bits within a wordline. Therefore, typically m is set equal to 8, 16, or 32. A linear expression of total memory array power without sense amplifiers, P array ,  405  is governed by
 
 P   array =( C   core   ·V   dd   2   ·f   op )+( m·V   dd   ·I   SA,bias )
 
Unlike the prior art, the dynamic power requirement here has two sets of linear traces. A first trace  405  relates to a reduced dynamic power requirement for a memory array without full sense amplifier operation and a second trace  407  relates to a reduced dynamic power requirement for a memory array with full sense amplifier operation. Both the first  405  and the second trace  407  dynamic power requirement occur prior to a critical frequency, f cr ,  409 . The critical frequency relates to a “slow mode” of memory cell operation and is inversely related to a critical access period, T cr , such that
 
f cr =T cr   −1  
 
The critical access period term T cr  will be developed shortly with reference to  FIGS. 5B and 5D , infra. Dynamic power requirements are reduced at frequencies less than f cr  due to a partial sense amplifier controlled power-down described in detail herein. At higher operational frequencies, that is, greater than f cr , the present invention operates with a dynamic power requirement similar to the prior art. Thus, a first high frequency operation trace  411  and a second high frequency operation trace  413  are indicative of dynamic power requirements without and with sense amplifier operation, respectively.
 
Automatic Address Transition Detection (ATD) Circuit
 
   With reference to  FIG. 5A , an exemplary embodiment  500  of the present invention includes an automatic ATD circuit  501 , a sense amplifier  503 , a sense amplifier bias circuit  505 , and a DQ flip-flop  507 . The embodiment is interspersed with other portions of a memory array as will be recognized by one of skill in the art. The other portions of the memory array are shown merely to provide a schematic relationship of the present invention to a typical memory array. 
   Below a “critical frequency,” f cr , the automatic ATD circuit  501  senses whenever an address change occurs and provides a sense amplifier enable, SA_EN, signal to activate sense amplifiers within the memory array. A user can change a system clock frequency over a large range but a system-clock-independent SA_EN signal is determined by the automatic ATD circuit  501  without operator intervention. The automatic ATD circuit  501  senses when an address (ADDR) signal transitions and sends a signal for the sense amplifier to turn on, allowing for time periods sufficient for ramp-up of current to the sense amplifier and charging of the sense amplifier lines (i.e., the sense amplifier is activated during a valid data out period. The critical frequency, f cr , is defined by particular characteristics within a given memory array circuit as explained in detail, infra. Above the critical frequency, the automatic ATD circuit  501  sends a constant SA_EN signal, allowing sense amplifiers to be constantly activated. 
   With reference to  FIG. 5B  an exemplary embodiment  501 A of the automatic ATD circuit  501  of  FIG. 5A  includes a delay circuit  511 , a rising-edge detection circuit  513 , a critical period delay element  515 , a first DQ flip-flop  517 , an optional DQ flip-flop  519 , and an OR gate  521 . 
   A skilled artisan will recognize that the delay circuit  511  may be constructed in various ways. For example, an appropriate delay may be achieved by constructing an even number of inverters is series; the higher a number of inverters placed in series, the greater the delay. The initial time delay is chosen to allow a comparison of the SYS_CLK to an output of the critical period delay element  515 , thus allowing any positive edge-triggered flip-flop to be used as a time comparator or phase detector. Therefore, if a signal input to the “D” input of the first DQ flip-flop  517  is “0” when the SYS_CLK goes high, then the system period is not short enough to disable SA_EN. Consequently, the automatic ATD detection circuit  501  remains in the “SLOW MODE” of operation ( FIG. 4 ). Details of exemplary embodiments of the rising-edge detection circuit  513  and the critical period delay element  515  are given in  FIGS. 5C and 5D , respectively. Determination of whether the optional DQ flip-flop  519  is included in the automatic ATD circuit  501 A will depend upon a range of SYS_CLK frequencies to which the circuit is subjected and overall stability considerations (e.g., when a period of the SYS_CLK is close to the critical period, T cr ). Such stability considerations are determinable by a skilled artisan (e.g., by circuit simulation). 
   Operation of the exemplary automatic ATD circuit  501 A is independent of a frequency of the system clock, SYS_CLK input. Instead, the exemplary automatic ATD circuit  501 A simply relies on the frequency of the SYS_CLK signal to determine when to produce a sense amplifier enable, SA_EN, signal and a duration of the signal. 
   Timing diagrams of  FIG. 5B  indicate a SA_EN signal for two different SYS_CLK frequencies, f 1  and f 2 . Recall that a period is simply an inverse of a clock frequency; thus 
             f   1     =     1     T   1             
The automatic address transition detection circuit  501 A compares frequencies of the SYS_CLK and an output of the critical period delay element  515 . An SA_EN signal is therefore produced only if a negative edge of the SAE_CR pulse (i.e., an output of the critical period delay element  515 ) occurs before a subsequent rising-edge of the SYS_CLK. T cr  is thus chosen to be longest period that will, overall, allow the sense amplifier  503  ( FIG. 5A ) to be on for the least amount of time possible, thereby saving power, but long enough in time to determine a memory cell state after an ATD signal occurs. Details of determination of the critical period, T cr , and relationships between the ADDR and SA_EN are developed with reference to  FIGS. 5D and 6 , infra.
 
Operation of the Automatic ATD Circuit (f 1 &lt;f cr )
 
   For a SYS_CLK frequency f 1 , a value of f 1  is such that T 1 &gt;T cr . In this case, a SYS_CLK signal, shown at “A,” is delayed, “B,” by the delay circuit  511 . A single pulse, at “C,” is produced as an output of the rising-edge detection circuit  513 . The single pulse at “C” is input to the critical period delay element  515 . A resultant pulse from the critical period delay element, at “D,” having a period T cr , produces a SAE_CR pulse which is one of at least two signal inputs to the OR gate  521 . (Details of the critical period delay element  515  are provided with reference to  FIG. 5D , infra.) The SYS_CLK initiates the pulse train at “D” and also provides an enable signal to the first DQ flip-flop  517  (as well as the optional DQ flip-flop  519  if present) on a rising-edge  523  of the f 1  SYS_CLK signal. Since the resultant pulse from the critical period delay element, at “D,” is low, a “0” is latched into the first DQ flip-flop  517 . As long as a period of the SYS_CLK is greater than T cr , (i.e., a frequency of the SYS_CLK is less than the critical frequency, f cr  (FIG.  4 )), then an SA_EN signal will only be produced when an address transition detection (ATD) occurs. 
   Operation of the Automatic ATD Circuit (f 2 &gt;f cr ) 
   For a SYS_CLK frequency f 2 , a value of f 2  is such that its related period T 2 &lt;T cr . In this case, a high frequency SYS_CLK signal, at “A,” is again delayed, shown at “B,” by the delay circuit  511 . As shown at “C,” a single pulse is produced as an output of the rising-edge detection circuit  513 . The single pulse at “C” is input to the critical period delay element  515 . The resultant pulse (i.e., the same pulse as describe supra with respect to the SYS_CLK frequency at f 1 ) from the critical period delay element, at “D,” having period T cr , produces a SAE_CR pulse which is input to the OR gate  521 . The SYS_CLK still initiates the pulse train at “D” and also provides an enable signal to the first DQ flip-flop  517  (and the optional DQ flip-flop  519 ) on a rising-edge  525  of the f 2  SYS_CLK signal. Here however, since the resultant pulse from the critical period delay element, at “D,” is high, a “1” is latched into the first DQ flip-flop  517 . Consequently an SA_EN signal appears high at an output of the OR gate  521 . 
   Operation of the Rising-Edge Detection Circuit 
   With reference to  FIG. 5C , an exemplary embodiment of a rising-edge detection circuit  513 A includes a first inverter  531 , a second inverter  533 , a PMOS transistor  535 , an NMOS transistor  537 , a third inverter  539 , and a AND gate  541 . Additionally included are analog components; a resistor having a value “r” and a capacitor having a value “c.” The rising-edge detection circuit  513 A is thus a hybrid analog-digital circuit. A combination of the PMOS transistor  535  and the NMOS transistor  537  essentially act as an inverter element. However, a combined effect of the resistor and capacitor produce a time constant, τ, such that a minimum time delay value, ∂ min , is the product of the resistive and capacitive values multiplied by the natural log value of “2.” Thus
 
∂ min   =rc ·[ln(2)]
 
where ∂ min  neglects minimal effects of digital component propagation delays. Consequently, any signal through the lower inverter leg portion of the rising-edge detection circuit  513 A will be further delayed in comparison to the signal traveling through the upper leg due to the lower leg analog components. For example, assuming a rising-edge appears at an input to the rising-edge detection circuit  513 A, a “fast 1” is produced at point “A.” After the first inverter  531 , a resulting “0” makes its way to the bottom leg, causing the PMOS transistor  535  to act as a pull-up device, creating a “1” as an input to the third inverter  539 . However, due to the delay going through the resistive and capacitive analog components, the signal is delayed by ∂ min  prior to passing through the third inverter  539 . At point “B,” a “slow 0” (or, otherwise put, a lingering “one”) is present due to the analog delay. Thus, a signal output from the AND gate  541  produces a narrow pulse only at times when both the top leg and bottom leg each are producing a high signal. A width, w, appropriate as an input to the critical period delay element  515  ( FIG. 5B ), may thus be chosen through proper selection of the resistive and capacitive elements.
 
Operation of the Critical Period Delay Element
 
   With reference to  FIG. 5D , an exemplary embodiment of a critical period delay element  515 A produces an output pulse having a width Δt based on the input pulse trigger from the rising-edge detection circuit  513 . A critical period, T c , is determined (for example, by circuit simulation) such that
 
 T   C   =t   on   +t   SA     —     delay   +t   off  
 
where t on , t SA     —     delay , and t off  will be defined with reference to  FIG. 6 , infra. The critical period delay element  515 A includes a PMOS transistor  551 , an NMOS transistor  553 , a resistor, a capacitor, and an inverter  555 . The critical period delay element  515 A functions similarly to the lower leg of the rising-edge detection circuit  513 A. Here, Δt=RC·[ln(2)], where “R” and “C” are resistive and capacitive values respectively that are chosen to give a pulse width Δt equal to the critical period, T C . The minimum width, w, of the input pulse output from the rising-edge detection circuit  513 A ( FIG. 5C ) is chosen to be long enough to fully discharge the capacitor. For this application, a value of the input pulse width, w, is typically less than 5 nanoseconds with an exemplary value of 3 nanoseconds minimizing total circuit delays. A trip point voltage, V tp , at point “A” sufficient to cause the inverter  555  to change states is simply
 
             V   tp     =       V   dd     2           
where V dd  is the system supply voltage.
 
Sense Amplifier Design
 
   With reference to  FIG. 5E , an exemplary sense amplifier  503 A is based on a transimpedance amplifier design, described in detail in U.S. Pat. No. 5,493,533, to Emil Lambrache (the inventor of the present invention described herein). U.S. Pat. No. 5,493,533 is hereby incorporated by reference in its entirety. 
   In brief, the sense amplifier  503 A is designed such that an output voltage, V out , is a function of a transimpedance transfer function, Z f , input current, I in , a reference current, I ref , and the supply voltage, V dd , according to the formula 
             V   out     =         Z   f     ⁡     (       -     I   in       +     I   ref       )       +       V   dd     2             
and V out  is a digital output voltage based on analog current inputs where I in =I cell  when reading a programmed memory cell such that
 
             V   out     =     {           0   ;               for   ⁢           ⁢     I   in       ≥       I   ref     ⁢           ⁢   when   ⁢           ⁢     I   in         =     I   cell                 V     dd   ;                 for   ⁢           ⁢     I   in       ≥       I   ref     ⁢           ⁢   when   ⁢           ⁢     I   in         =     0   ⁢     (     erased   ⁢           ⁢   cell     )                       
Further design considerations include determining a transimpedance transfer function, Z f , such that
 
               Z   ref     ·     I   ref       =       V   dd     2           
and determining a reference current I ref  such that
 
               I   ref     =       I   cell     2       ,         
where I cell ≅10 μA for a typical programmed memory cell.
 
Representative Timing Diagrams
 
   With reference to  FIG. 6 , an exemplary timing diagram of the automatic ATD circuit of  FIG. 5A  has a sense amplifier current graph, I SA  that begins to ramp up to a steady sense amplifier bias current, I SA     —     bias , as soon as a read enable, READ_EN, signal is asserted. Depending upon characteristics of the sense amplifier circuit, the sense amplifier bias current achieves a steady-state condition typically within 1 μs–10 μs. An address transition signal, ADDR[n:1], may be asserted after the SYS_CLK goes high. Upon detection of an ADDR[n:1] signal, the automatic ATD control circuit  501  ( FIG. 5A ) sends an SA_EN signal to the sense amplifier  503 , causing charge to be pumped into the sense amplifier (i.e., charge capacitor gate-to-source, C g-s  to pump an electron charge into a channel of the transistor). A time delay, t on , occurs while the sense amplifier is being charged. There is an additional delay, t SA     —     delay , that occurs while sense amplifier lines to the memory cell are charging. After the lines are fully charged, an SA_CLK signal is asserted, allowing a data output, D out , to be latched into the DQ flip-flop  507  ( FIG. 5A ). D out  will be valid until the SA_EN signal goes low, forcing the sense amplifier current to return to an I SA     —     bias  condition after a slight delay period, t off , where charge is bled off. Thus, a significant power savings may be realized by employing the automatic ATD control circuit. For example, if a low speed SYS_CLK operation has a frequency of 1 MHz and an SA_EN signal of 100 nsec is sufficient to enable data from a memory cell, then only 
               100   ⁢           ⁢   n   ⁢           ⁢   sec       1   ⁢           ⁢   μsec       =     10   ⁢   %           
of the power required to keep a sense amplifier at full power constantly is utilized by adoption of the present invention. Therefore, the critical time period, T c , noted above with regard to  FIG. 5D  (recall that T c =t on +t SA     —     delay +t off ) is calculated based on the delay times referenced in the I SA  graph.
 
   Note further that as the SYS_CLK frequency increases to a frequency slightly greater than t SA     —     delay , there is no advantage in turning the sense amplifier off as an inherent charge wasted (indicated by an integration of the hatched areas “A” representing charge pumped in during t on  and charge bled off during t off ) is greater than any possible energy savings. However, the SYS_CLK frequency may be constantly changed and the automatic ATD control circuit  501  will determine an optimal timing determination for turning the sense amplifier on or off or leaving the sense amplifier on constantly. This automatic determination feature is exemplified with reference to  FIGS. 5B and 7 . 
     FIG. 7  indicates a SYS_CLK at a first frequency until point “A” whereupon the SYS_CLK switches to a second frequency. At point “B” the SYS_CLK changes to a third frequency (or back to the first frequency). An EDGE DETECT graph indicates an output from the rising-edge detection circuit  513  (point “C” in  FIG. 5B ). Note that both a width of the edge detect pulse and an SAE_CR signal are constant despite a change in the SYS_CLK frequency. For example, note that at a rising-edge of the SYS_CLK at a first rising-edge time  703   1 , an output of the critical period delay element  515  (point “D” in  FIG. 5B ) goes high and returns to “0” prior to the next rising-edge of the SYS_CLK pulse at a second rising-edge time  703   2 . Thus the circuit is operating in a low power operation mode, keeping the sense amplifier in a low power mode (i.e., “SLOW MODE,”  FIG. 4 ) by supplying operational current to the sense amplifier only as long as needed. However, at point “B” where the SYS_CLK frequency changes to a frequency greater than the critical frequency, f cr , ( FIG. 4 ) such that the SAE_CR pulse is unable to return to “0” prior to a subsequent rising clock edge of the SYS_CLK at a third rising-edge time  7033 . Thus, the output of the first flip-flop  517  which is an ATD_DISABLE signal is asserted at the third rising-edge time  703   3  and remains high during a period of high frequency operation  705  continuing through subsequent rising-edge times  703   4 ,  703   5 . The optional DQ flip-flop  519  creates a second ATD_DISABLE signal in case the first one has a glitch when f is approximately equal to f cr  and the first DQ flip-flop  517  is left in a metastable state. During the second low frequency operation period  707 , the automatic ATD control circuit  501  restarts the low power operation mode beginning at a fourth rising-edge time  703   6 . 
   In the foregoing specification, the present invention has been described with reference to specific embodiments thereof. It will, however, be evident to a skilled artisan that various modifications and changes can be made thereto without departing from the broader spirit and scope of the invention as set forth in the appended claims. For example, skilled artisans will appreciate that the DQ flip-flops of  FIG. 5B  may be substituted with other components to achieve a similar time comparison function. For example, a Schmitt trigger could be used in place of the optional DQ flip-flop  519  of  FIG. 5B . Further, other circuits may be substituted for the rising-edge detection circuit  513 A and the critical period delay element  515 A of  FIGS. 5C and 5D  respectively. Further, the rising-edge detection circuit may be reconfigured, with appropriate timing considerations, to operate on a falling-edge of the clock. The resistors and capacitors described herein may similarly be substituted by appropriate resistive and capacitive elements as known in the art. The specification and drawings are, accordingly, to be regarded in an illustrative rather than a restrictive sense.