Abstract:
A dual-polarized radiating element is formed from two orthogonally oriented monopole radiators disposed on a dielectric substrate. An RF image plane placed orthogonally to the two monopole radiators presents a balanced excitation for element impedance optimization that allows for operation over multiple octave bandwidths with a physically compact device. The dual-polarized radiating element provides a broad field-of-view (FOV) as a stand alone radiator and may be used in a phased array.

Description:
FIELD OF INVENTION 
     The present invention relates, in general, to an antenna and, more specifically, to a compact radiating element that may be deployed as a single radiator or configured for use in a phased array. The radiating element operates over multioctave bandwidths, subtends a wide field-of-view (FOV), and responds to any desired polarization in space. The present invention may operate at high peak and average power in the transmit mode and is amenable to conformal installation. 
     BACKGROUND OF THE INVENTION 
     It is well known that the efficiency of an antenna diminishes significantly as its dimensions decrease to much less than a wavelength. In such instances complex tuning networks are employed to match the antenna radiation resistance to the transmitter or receiver, where the major portion of the signal is dissipated in the matching network. For example, airborne towel-bar blades operating at VHF/UHF frequencies may exhibit gains as low as −30 dBiL in the lower segments of the operating band. Apostolos in U.S. Pat. No. 5,790,080, entitled “Meander Line Loaded Antenna”, discloses that an antenna design may be conceived on a volumetric basis rather than a planar basis, where the limitation on performance is governed by the well known Chu-Harrington relationship that allows an antenna aperture to be much less than a wavelength in its operating frequency band. 
     In vehicular or airborne applications where space is at a premium and there is a need for efficient antennas operating in the VHF/UHF bands, volumetric solutions to antenna problems are imperative. Additionally, modern systems employ polarization as a significant parameter during system processing and transmission. Consequently, not only must the antenna be compact, but it must also provide independent orthogonal linearly-polarized components to avail the system processors of polarization diversity. 
     A figure of merit for providing an efficient radiating element is the net gain expressed by the familiar relationship:
 
G=ηD
         where: G is the net gain of the antenna
           η is the antenna efficiency, and   D is the antenna directivity
 
The directivity of a radiator may be defined by the radiated beamwidth of the antenna:
 
 D= 4π/θφ
   
           where: θ and φ are half-power beamwidths expressed in radians.       

     With the directivity established by the beamwidths of the radiated element patterns, which cover a broad field-of-view and are reasonably stable with frequency, the improvement in antenna gain may only be achieved by maximizing the antenna efficiency η. In practice, this translates into optimizing the antenna input VSWR, the voltage standing wave ratio, over the operating bandwidth and employing elements with minimum insertion loss. 
     This present invention addresses the needs enumerated above, as well as other needs, such as radiating high pulsed and CW power during transmission. 
     The present invention is related to U.S. Pat. No. 6,853,351, entitled “Compact High-Power Reflective-Cavity Backed Spiral Antenna” by Mohuchy, and U.S. Pat. No. 7,372,424, entitled “High Power, Polarization-Diverse Cloverleaf Phased Array”, also by Mohuchy, issued on May 13, 2008, the contents of which are hereby incorporated by reference in their entireties. 
     SUMMARY OF THE INVENTION 
     A radio frequency (RF) transmitting and receiving device constructed in accordance with the present invention provides a compact, broadband radiating element with two independent orthogonally-polarized field components. The radiating element includes two radiating microstrip surfaces disposed conformally on a planar substrate in a butterfly-wing arrangement. Each radiating microstrip surface includes an RF launch point and an orthogonal metallic strip for optimizing the input VSWR. Each radiating surface extends beyond and folds over an edge of the radiating element in a predetermined manner which is configured to extend performance at the low end of the operating frequency band. The radiating microstrips of the present invention are disposed at a distance that is less than one-quarter wavelength above a metallic ground plane. 
     The present invention includes an imaging surface in proximity to the RF launch point of each radiating element. The imaging surface is oriented orthogonally to the metallic ground plane. In this manner, each monopole behaves electrically as a dipole in terms of gain, radiation pattern and input VSWR, and uses only half of the surface area. 
     It is understood that the foregoing general description and the following detailed description are exemplary, but are not restrictive of the invention. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWING 
       The invention may be best understood from the following detailed description, when read in conjunction with the accompanying drawings. Included within are the following figures: 
         FIG. 1  is a perspective view of the inventive monopole radiating antenna element configured in a triangular butterfly pattern that is conformally mounted as microstrips on a multilayer substrate to form a planar radiating surface, according to an embodiment of the present invention. 
         FIGS. 2A ,  2 B and  2 C are different views of the monopole radiating antenna element shown in  FIG. 1 , according to an embodiment of the present invention. 
         FIG. 3  is a schematic view of the monopole radiating antenna element shown in  FIG. 1 , depicting two radiating surfaces and two imaging radiating surfaces, according to an embodiment of the present invention. 
         FIG. 4  is a view of the RF feed attachment to the monopole radiating antenna element shown in  FIG. 1 , according to an embodiment of the present invention (only a portion of the monopole radiating antenna element is shown). 
         FIG. 5  depicts an RF conductor included in the feed arrangement of the monopole radiating antenna element shown in  FIG. 1 , according to an embodiment of the present invention. 
         FIG. 6  is a plot of input return loss versus frequency of an exemplary monopole radiating antenna element shown in  FIG. 1 , according to an embodiment of the present invention. 
         FIG. 7  is a plot of gain versus frequency of an exemplary monopole radiating antenna element shown in  FIG. 1 , according to an embodiment of the present invention. 
         FIG. 8  is a top view of a butterfly arrangement of two radiating surfaces of the monopole radiating antenna element, in accordance with an embodiment of the present invention. 
         FIG. 9  is top view of another butterfly arrangement of two radiating surfaces of the monopole radiating antenna element, in accordance with another embodiment of the present invention. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Referring to  FIG. 1 , there is shown a perspective view of the monopole radiating antenna element, in accordance with an embodiment of the present invention. As shown, the monopole radiating antenna element, designated as  4 , includes two radiating surfaces  6  (also referred to herein as radiating elements  6 ), which are arranged as an orthogonal pair in a butterfly pattern. The orthogonal pair of radiating elements is formed conformally on a thin substrate  11  and is oriented at 45° with respect to a principal antenna element axis, designated as  3 . The two radiating surfaces, which are arranged in an X,Y plane, extend beyond their surface dimensions, as they are folded into an X,Z plane (shown as two fold-over extensions  8 ). 
     The substrate  11  is mounted on a layer of dielectric material, designated as  12 . The dielectric layer  12  is supported by a reflective metallic ground plane, designated as  13  (disposed in an X,Y plane). An RF imaging plane (also disposed in an X,Y plane) is formed by metallic surface  10  (the latter disposed in a Y,Z plane). As will be explained below, the RF imaging plane is oriented perpendicular to RF launchers  7 . The metallic surface  10  is separated from the two radiating surfaces  6  by an electrically determined separation distance X (shown best in  FIG. 2A ). In addition, it will be appreciated that the RF imaging surfaces (shown as  6   a  in  FIG. 3 ) are separated from the two radiating surfaces  6  by an electrically determined separation distance  2 X. 
     The monopole radiating antenna element is shown in more detail in  FIGS. 2A ,  2 B and  2 C.  FIG. 2A  is a perspective view of the monopole radiating antenna element  4 , showing the perpendicular orientation between the two radiating elements  6  and metallic surface  10 . Also shown are RF conductors  14  that extend in a generally parallel direction to metallic surface  10  and meet RF launchers  7  ( FIG. 2B ) of radiating surfaces  6  in a generally perpendicular direction. 
       FIG. 2B  shows the two radiating surfaces (or elements)  6  of the monopole radiating antenna element  4 . Also shown are the two fold-over extensions  8  that are oriented perpendicularly to elements  6 . Each fold-over extends, as shown, by a distance of D. Also shown are the two RF launchers  7  positioned adjacent the distal ends of radiating surfaces  6  and near metallic surface  10 . The RF launchers  7  also intersect orthogonal lines  5  (shown in  FIG. 1 ). 
     When the inventive radiating elements  6  are deployed in a phased array configuration, the fold-over extension  8  may be eliminated. The inter-element mutual coupling may then be employed to provide desired broadbanding effects. 
     The RF signal is inputted, or received by a transmission medium, such as RF conductors  14 , shown in a perspective view in  FIG. 2A . Each RF conductor  14  connects RF terminal  16 , shown in  FIG. 5 , with a respective launcher  7 . The RF conductors  14  may also be employed as an impedance transformer between a 50 ohm coaxial input at RF terminals  16  and the radiating elements  6 . The choice of a 50 ohm input may be based on the impedance of the transmission line and may be varied to accommodate any input transmission line. In such case, the impedance of transformer  14  (or RF conductors  14 ) may be selected appropriately. At the output of each RF conductor  14 , there may be included a capacitive metallic strip, designated as  15 , in order to provide additional impedance tuning and extend the useful bandwidth of the inventive radiating antenna element  4 . As shown in  FIG. 2C , RF conductor  14  is electrically connected to capacitive metallic strip  15 . 
     It will be appreciated that radiating elements  6  in  FIG. 2B  may be formed to occupy the maximum available surface area of the top surface of substrate  11 , except for the tapers near each RF launcher  7 . The tapers may be determined empirically for a minimum input VSWR, using methods well established in the art. Additionally, fold-over extensions  8  may also be determined empirically, while focusing on extending performance at the low frequencies. 
     A performance tradeoff may be done to determine the distance D of fold-over extensions  8  and their interaction with ground plane  13  (as best shown in  FIG. 1 , ground plane  13  is disposed substantially parallel to substrate  11  with dielectric layer  12  sandwiched in-between). 
     Similarly, the dimensions of capacitive metallic strip  15  may be determined empirically for the best input VSWR. The dimensions of capacitive metallic strip  15  are shown in  FIG. 2C , as having length A and height B. 
     Other methods known in the art may be employed to perform RF tuning functions, such as tuning with tank circuits, but they are more complex and result in a decrease of radiator efficiency. 
     The RF imaging surfaces will now be described by reference to  FIG. 3 . As shown, metallic surface  10  forms an RF imaging plane of the present invention. The RF imaging plane, which is formed in the same plane as radiating surfaces  6 , are disposed adjacent to RF launchers  7  and perpendicular to RF conductors  14 . The close placement of RF launchers  7  to metallic surface  10  effectively forms an electrical simulation of radiating surfaces  6   a  and  8   a . The simulated radiating surfaces  6   a  and  8   a  are mirror images of radiating surfaces  6  and  8 , respectively. As described above, the two simulated radiating surfaces  6   a  are separated from the two radiating surfaces  6  by an electrically determined separation distance  2 X. The metallic surface  10  extends between the simulated radiating surfaces  6   a  and radiating surfaces  6 . 
     From an input impedance perspective, the stimulated radiating surfaces  6   a  and  8   a  represent a balanced line excitation of each monopole  6  and expand the useful bandwidth of the present invention. In effect, each monopole  6  exhibits radiation characteristics of a broadband dipole. 
     Still referring to  FIG. 3 , the polarization diversity of the present invention will now be described. The invention may be configured to achieve full polarization diversity with the present monopole radiator. Using the left monopole  6  as a reference with an electric field excitation E, as shown in  FIG. 3 , if the right monopole  6  is excited with E 1  at a phase angle φ 1  set to zero degrees and the left monopole  6  is excited with E, the resultant radiated field is linearly polarized in the X direction. Conversely, if the right monopole  6  is excited with E 2  at a phase angle φ 2  set to zero degrees and the left monopole  6  is excited with E, the resultant radiated field is linearly polarized in the Y direction. 
     A full complement of linear polarizations in the X,Y plane may be realized by varying the excitation amplitudes of the relative field strengths. Circular polarization may be realized by setting the field phase angles φ n  to +90° or −90° for either right hand circular radiation or left hand circular radiation. Any elliptical polarization may result by varying the phase angles φ n . 
     The radiating elements  6  may be formed by chemically etching the copper clad dielectric material of substrate  11 . The radiating elements  6  are shown in  FIGS. 1 ,  2 A,  2 B,  3  and  4  ( FIG. 4  shows a portion of radiating elements  6 ). 
     Connectivity to each of the RF conductors  14  may be achieved using flat socket screws  20  to provide good electrical contacts to respective launchers  7  of radiating elements  6 , as shown in  FIGS. 4 and 5 . Solid metallic plates  21  may be included between the etched radiating elements  6  and screws  20  to assure that radiating elements  6  remain in place during the attachment process. 
     A transmission line, generally designated as  21 , as shown in  FIG. 5 , includes coaxial bulkhead connector  16  with its dielectric sleeve  18  extending a distance T. The distance T is determined by the thickness of ground plane  13 , which is disposed at the bottom of monopole radiating antenna element  4 , as shown in  FIG. 1 . The center conductor of each coaxial connector  16  is positively joined to a respective RF conductor  14  with set screw  19 . 
     The RF conductors  14  for the radiating elements  6  may be arranged as a balanced twin-lead transmission line pair in conjunction with simulated radiating surfaces  6   a  formed by image plane  10 . The socket set screw  20  caps an end of RF conductor  14  to provide a positive connection to each radiating surface  6 , thereby adding mechanical integrity. Also shown is flange  17  for providing a sturdy connection to ground plane  13  by way of screws (not shown) inserted through flange  17  and ground plane  13 . 
     An exemplary monopole radiating antenna element  4  was fabricated and measured in the 100-800 MHz frequency band. A baseline for the monopole radiating aperture was determined using the general guidelines for biconical antennas as outlined by J. D. Kraus in “Antennas”, second edition, published by McGraw-Hill Book Co, 1988, chapter 2. The initial dimensions were then optimized using a three-dimensional Finite Element Analysis (FEA) tool that allows construction of the monopole elements. Exemplary radiation patterns and driving port impedances were computed using numerical computation techniques and accounting for the contributions of the radiating surface extensions and the reactance at the input of the radiating antenna element. 
     The dimensions of the exemplary antenna were optimized for a maximum operating bandwidth centered at 350 MHz. The tradeoff parameters in  FIGS. 2A ,  2 B and  2 C were antenna element volume defined by the length L, the width W and the depth H. From a network point of view, the length L behaves as an inductive component, while the width W and the height of the fold-over extensions D represent capacitance. Additional capacitance may be obtained by varying length A of metallic strip  15  from the element feed points (RF launchers  7 ). The combined effect provides a tank circuit which may be optimized for maximum operating bandwidth. 
     A good performance indicator of the radiating antenna element is the VSWR (Voltage Standing Wave Ratio) for both the input to the antenna element from the RF feed and the return loss seen by an incoming plane wave into the antenna element. A desired figure of merit for both conditions may be to operate a broadband antenna element with a VSWR under 2:1. In practice, however, operating an antenna element up to a VSWR of 3:1 ratio may be used, without significantly degrading the overall operating efficiency. It will be appreciated that although this remains a practical bound for high power applications, even wider bandwidths may be possible for low power transmissions or receptions. 
       FIG. 6  shows an optimized VSWR performance for the present invention when measured at the coaxial TNC input connector, whose characteristic impedance is 50 ohms. The designation V represents an E-field orientation in the X axis and the designation H represents an E-Field orientation in the Y axis. 
     A relationship between VSWR and return loss in  FIG. 6  may be expressed as follows:
 
ρ=(σ−1)/(σ+1)
         Where: ρ is Return Loss in voltage ratio
           σ is VSWR in voltage ratio.   
               

     Exemplary dimensions derived from the optimization may be:
         L=22.4 inches   W=11.0 inches   H=7.22 inches       

     The fold-over extensions D may be 2.4 inches. 
     The length A of the metallic strips from the feed point may be 3.0 inches. 
     The dielectric constant of the material of substrate  12  may be 1.35. 
     It will be understood that when the dielectric constant of the substrate is changed, the depth H of the antenna element may also be adjusted using techniques well known in the art. 
     The center RF conductors of transmission lines  21  (only one is shown in  FIG. 5 ), behave electrically as described and shown as RF conductors  42  and  43  in FIG. 4 of U.S. Pat. No. 6,853,351, which is incorporated herein by reference. The impedance, and hence the dimensions of the center RF conductors may be determined by appreciating that they form a pair of transmission lines connecting the input of the antenna element to the individual radiating elements. The center RF conductors may also be approximately λ/4 long, an ideal electrical length for a quarter-wave transformer. The calculated impedance at the feed points of each radiating element is 160 ohms. The RF connectors, when disposed in the presence of the image plane, effectively represent 100 ohms. The resultant impedance then becomes 126 ohms, which corresponds to a conductor diameter of 0.34 inches. 
     The measured gain of the exemplary antenna element to matched polarization is shown in  FIG. 7 . While these measurements were performed in an anechoic chamber equipped to operate from 200 MHz through 500 MHz, the useful antenna bandwidth is shown in  FIG. 6 . 
     Another embodiment of the present invention is shown in  FIG. 8 , where a top view of two radiating surfaces  82  are illustrated. Both radiating surfaces are arranged in the X,Y plane on substrate  86 , and extend into the X,Z plane, as fold-over extensions  8 . Similar to the embodiment shown in  FIG. 1 , radiating surfaces  82  are arranged as an orthogonal pair in a butterfly pattern. The orthogonal pair is formed conformally on substrate  86  and oriented at 45° with respect to principal antenna axis  3 . Two orthogonal lines  5  intersect, as shown, the principal antenna axis. 
     Proximate to principal antenna axis  3 , each radiating surface  82  forms two perpendicular edges extending in the X and Y directions, away from the origin point of the X, Y, Z axes. Adjacent to each intersection of the two perpendicular edges, an RF launcher, designated as  84 , extends in the Z direction, perpendicular to substrate  86 . The RF launchers  84  also intersect the two orthogonal lines  5 . 
     As shown, each of the two orthogonal lines  5  intersects (a) two perpendicular edges proximate to an RF launcher  84  and (b) two perpendicular edges formed distally on substrate  86  by a respective radiating surface  82 . The one edge in the Y direction, proximate to RF launcher  84 , has a clearance of ΔX away from the end of substrate  86 . There is a separation of 2ΔX between the other edges in the X direction of the two radiating surfaces  82 . 
     Extending between (a) the two perpendicular edges proximate to RF launcher  84  and (b) the two perpendicular edges disposed distally from RF launcher  84  are respective edges  87  and  89  of each radiating surface  82 . The edge  87  makes an angle of 20° (for example, as shown) with respect to the Y axis. The edge  89  makes an angle of 25° (for example, as shown) with respect to the X axis. 
     A notch, as shown in  FIG. 8 , is formed between each edge  87  and one of the two perpendicular edges formed distally from each RF launcher  84 . The notch has a width of ΔY. The fold-over extensions into the Z axes (best illustrated in  FIG. 2B ) are shown designated as  8 . 
     Another embodiment of the present invention is shown in  FIG. 9 , where a top view of two radiating surfaces  92  are illustrated. Both radiating surfaces are arranged in the X,Y plane on substrate  96 , and extend into the X,Z plane, as fold-over extensions  8 . Similar to the embodiment shown in  FIG. 8 , radiating surfaces  92  are arranged as an orthogonal pair in a butterfly pattern. The orthogonal pair is formed conformally on substrate  96  and oriented at 45° with respect to principal antenna axis  3 . Two orthogonal lines  5  intersect, as shown, principal antenna axis  3 . 
     Proximate to principal antenna axis  3 , each radiating surface  92  forms two perpendicular edges extending in the X and Y directions, away from the origin point of the X, Y, Z axes. Adjacent to each intersection of the two perpendicular edges, an RF launcher, designated as  94 , extends in the Z direction, perpendicular to substrate  96 . The RF launchers  94  also intersect the two orthogonal lines  5 . 
     As shown, each of the two orthogonal lines  5  intersects (a) two perpendicular edges proximate to an RF launcher  94  and (b) two perpendicular edges formed distally on substrate  96  by a respective radiating surface  92 . The one edge in the Y direction, proximate to RF launcher  94 , has a clearance of ΔX away from the end of substrate  96 . There is a separation of 2ΔX between the other edges in the X direction of the two radiating surfaces  92 . 
     Extending between (a) the two perpendicular edges proximate to RF launcher  94  and (b) the two perpendicular edges disposed distally from RF launcher  94  are respective edges  97  and  99  of each radiating surface  92 . The edge  97  makes an angle of 20° (for example, as shown) with respect to the Y axis. The edge  99  makes an angle of 25° (for example, as shown) with respect to the X axis. 
     It will be appreciated that the notch shown in  FIG. 8  with a width of ΔY is missing in  FIG. 9 , as ΔY equals zero in  FIG. 9 . The fold-over extensions  8  into the Z axis extend along the entire lengths of the perpendicular edges in the X direction formed at the ends of the substrate surface. 
     Having described an exemplary embodiment of this invention, it is evident that other embodiments incorporating these concepts may be used. For example, frequency scaling of the dimensions may be used to operate in other frequency bands. The types of fasteners, connectors or dielectrics may be varied, with the appropriate electrical compensation. The antenna element may be used in a planar or a conformally shaped phased array structure deployed to any aspect ratio commensurate with the intended spatial coverage. In such applications, the fold-over extension may be excluded and replaced by mutual coupling between adjacent radiating elements, as described in U.S. Pat. No. 7,372,424, which is incorporated herein by reference. 
     Accordingly, although the invention has been described in one exemplary form with a certain degree of particularity, it is understood that the present disclosure is made only by way of example and that numerous changes in the details of construction and combination of parts may be made without departing from the spirit and the scope of the invention.