Abstract:
A capacitance-to-digital converter for an extended range of capacitances includes a reference capacitor and one or more offset capacitors. Electrical charge accumulated in the offset capacitors is used to at least partially cancel the charge accumulated in a sensed capacitance to facilitate matching with a charge accumulated in the reference capacitor. The residual charge is passed to an integrator, the output from which is quantized and used to control switching of the capacitors. Immunity to tonal external noises and improved conversion speed are achieved by controlling the capacitor switching with a spread spectrum clock. The capacitance-to-digital converter may be used, for example, for sensing of the capacitances of capacitive elements in touch and proximity displays or other user interfaces.

Description:
BACKGROUND 
     The capacitance of a capacitive sensor changes when an object approaches or touches the sensor. Since the sensors require no moving parts, capacitive sensors may be robust and reliable and widely used in many areas. In particular, capacitive sensors are used in human-to-machine interfaces such as buttons, jog wheels, switches, scroll bars and touch screens. 
     In many applications, capacitive sensors interface to digital electronic controllers via a capacitance-to-digital converter. Sigma-delta capacitance-to-digital converters have been used successfully in many applications. In a sigma-delta converter, a sigma delta modulator generates a binary sequence of zeros and ones that indicate whether the charge accumulated by the capacitance of the sensor is greater than or less than a reference charge accumulated on a reference capacitor. The sequence of zeros and ones may be integrated and decimated to determine the relationship of the sensor&#39;s capacitance to the reference capacitance. 
     One limitation of this approach is that the reference capacitance must be greater than the sensor capacitance. However, if the capacitance is too large, the sensitivity of the converter is reduced. One approach to reduce this limitation is to adjust the sampling time of the reference capacitance relative to the sampling time of the sensor capacitance. Another approach is to a use an additional offset capacitor that is clocked out of phase with the excitation signal. A still further approach is to adjust the voltage of the excitation signal. 
     In practice, the impedance of the sensor is not purely capacitive. Hence, a further limitation is that the conversion is that the conversion speed is limited by the discharge time of the sensor capacitance. The discharge time increases as the resistive component of the sensor impedance increases. This can be a significant limitation for applications such as touch screens, which utilize a matrix of sensing elements and require multiple conversions for a single position estimate. 
     A further limitation is that electromagnetic interference generated by the converters is concentrated in very narrow frequency bands that are multiples of the clock frequencies. 
     A still further limitation is that a converter may be sensitive to noise, such as electromagnetic interference from synchronous components. 
    
    
     
       BRIEF DESCRIPTION OF THE FIGURES 
       The accompanying figures, in which like reference numerals refer to identical or functionally similar elements throughout the separate views and which together with the detailed description below are incorporated in and form part of the specification, serve to further illustrate various embodiments and to explain various principles and advantages all in accordance with the present invention. 
         FIG. 1  is an example capacitance-to-digital converter in accordance with some embodiments of the invention. 
         FIG. 2  is an example a switched offset capacitance in accordance with some embodiments of the invention. 
         FIG. 3  is a graphical representation of an exemplary charge transfer function. 
         FIG. 4  is a graphical representation of clock signals and an integrator voltage in a capacitance-to-digital converter. 
         FIG. 5  is a graphical representation of clock signals in a capacitance-to-digital converter in accordance with some embodiments of the invention. 
         FIG. 6  and  FIG. 7  are graphical representations of clock signals and integrator voltages in a capacitance-to-digital converter in accordance with some embodiments of the invention. 
         FIG. 8  is a simplified block diagram of a touch and proximity sensor in accordance with some embodiments of the invention. 
         FIG. 9  is further block diagram of a touch and proximity sensor in accordance with some embodiments of the invention. 
         FIG. 10  is a graphical representation of clock signals generated by a random or pseudo-random sequence in a capacitance-to-digital converter, in accordance with some embodiments of the invention 
     
    
    
     Skilled artisans will appreciate that elements in the figures are illustrated for simplicity and clarity and have not necessarily been drawn to scale. For example, the dimensions of some of the elements in the figures may be exaggerated relative to other elements to help to improve understanding of embodiments of the present invention. 
     DETAILED DESCRIPTION 
     Before describing in detail embodiments that are in accordance with the present invention, it should be observed that the embodiments reside primarily in combinations of method steps and apparatus components related to capacitance testing. Accordingly, the apparatus components and method steps have been represented where appropriate by conventional symbols in the drawings, showing only those specific details that are pertinent to understanding the embodiments of the present invention so as not to obscure the disclosure with details that will be readily apparent to those of ordinary skill in the art having the benefit of the description herein. 
     In this document, relational terms such as first and second, top and bottom, and the like may be used solely to distinguish one entity or action from another entity or action without necessarily requiring or implying any actual such relationship or order between such entities or actions. The terms “comprises,” “comprising,” or any other variation thereof, are intended to cover a non-exclusive inclusion, such that a process, method, article, or apparatus that comprises a list of elements does not include only those elements but may include other elements not expressly listed or inherent to such process, method, article, or apparatus. An element preceded by “comprises . . . a” does not, without more constraints, preclude the existence of additional identical elements in the process, method, article, or apparatus that comprises the element. 
     It will be appreciated that embodiments of the invention described herein may a one or more application specific integrated circuits (ASICs), in which each function or some combinations of certain of the functions are implemented as custom logic. Thus, methods and means for these functions have been described herein. Further, it is expected that one of ordinary skill, notwithstanding possibly significant effort and many design choices motivated by, for example, available time, current technology, and economic considerations, when guided by the concepts and principles disclosed herein will be readily capable of generating such integrated circuits with minimal experimentation. 
       FIG. 1  is an example capacitance-to-digital converter in accordance with some embodiments of the invention. In  FIG. 1 , the converter  100  generates a binary sequence of zeros and ones  102  that indicate whether the charge accumulated by the capacitive element  104  of a sensor  106  is greater than or less than a reference charge generated by a reference charge generator  107 . In the embodiment shown in  FIG. 1 , the reference charge generator  107  includes a reference capacitive element  108 . The binary sequence  102  may be integrated and decimated in a filter  110 , which may be integrated in the same circuit as the converter  100 , to determine the relationship of the sensor&#39;s capacitive element  104  to the reference capacitive element  108 . The capacitive element  104  of the sensor  106  is driven by an excitation voltage  112  that is output from the converter  100  and produces an output signal  114  that is fed back to the converter  100 . In this example, the excitation voltage  112  is provided by a voltage reference source  116  that is coupled via a clocked transmission gate  118 . The gate  118  is driven by a first clock signal, denoted as Φ 1 . During a sampling interval when the first clock signal is asserted, the gate  118  is closed and the capacitance of the sensor samples the excitation voltage. The capacitive element  104  is also coupled to ground via another clocked transmission gate  120  driven by a second clock signal, denoted as Φ 2 . In contrast to prior converters, the signal Φ 2  is not the complement of Φ 1 , as will be discussed in more detail below. 
     The reference capacitive element  108  of the reference capacitance generator  107  is coupled to the voltage reference source  116  by a clocked transmission gate  122  and to ground via a clocked transmission gate  124 . During an integration interval, the clocked transmission gate  126  is closed and the capacitances  104  and  108  are coupled to an integrator  128 . In the embodiment shown, the integrator  128  comprises a capacitor ‘Cint’ coupled around an amplifier ‘A’. When the clocked transmission gate  130  is closed the capacitive elements are coupled to ground. If gate  122  is closed during the sampling interval, the reference capacitive element  108  also samples the reference voltage, and the sum of the accumulated charges is fed to the integrator  128 . If gate  122  is closed during the integration interval, charge is held in the reference capacitive element  108  and the difference of the charges is passed to the integrator. Thus, the relative timing of the transmission gates controls the charge fed into the integrator  128 . The output from the integrator  128  is passed to a comparator  132  that compares the integrated value to a threshold (such as ground, for example) and generates the binary output sequence  102 . A clock generator  134  generates the first and second signal clock signals  136 . The clock generator may incorporate a pseudo random noise sequence (PRNS) generator, as will be discussed below with reference to  FIG. 10 . The binary output sequence  102  is input to feedback logic  138  where it is combined with the first and second clock signals Φ 1  and Φ 2  to produce the switch control signals  140  (Φ 3  and Φ 4 ) that control the clocked transmission gates of the reference capacitance generator  107 . If the integrator output is high, the gates  118  and  122  are operated out of phase (e.g. Φ 3 =Φ 2 , Φ 4 =Φ 1 ). If the integrator output is low, the gates  118  and  122  are operated in phase (e.g. Φ 3 =Φ 1 , Φ 4 =Φ 2 ). 
     In general, control of the clocked transmission gates  118 ,  120 ,  126  and  130  is independent of the comparator output  102 , while control of the gates  122  and  124  is dependent upon the comparator output. 
     In one embodiment, the dynamic range of the converted is increased by adding a charge compensation circuit  142 . The charge compensation circuit  142  generates a charge  144  that at least partially compensates for the charge accumulated in the sensor capacitive element  104 . In the exemplary embodiment shown in  FIG. 1 . An offset capacitor  146  samples a reference voltage when clocked transmission gate  148  is closed and is coupled to the integrator  128  when clocked transmission gate  150  is closed. The sampling is synchronized with the sampling of the sensor capacitive element  104 . 
     The offset capacitor may be a selectable capacitor, the capacitance of which is controlled by a signal on control line  152 . The control signal is generated from an offset select logic circuit  154 . In one embodiment the capacitance of the selectable offset capacitor is selected dependent upon the binary output sequence  102 . For example, the capacitance may be selected dependent upon the ratio of ones to zeros in the binary output sequence. In a further embodiment, the capacitance is programmed dependent upon characteristics of the sensor  106 . 
     In practice, the sensor  106  is not purely capacitive, but includes a resistive component, as indicated by the series resistors  156  and  158  shown  FIG. 1 . The series resistance indicated by the series resistors  156  and  158  slows the rate at which the capacitive element  104  can be charged or discharged. This will be discussed below with reference to  FIG. 3 . 
       FIG. 2  shows an exemplary offset capacitor  146 . The capacitor  146  has a selectable capacitance and comprises a number of individual capacitive elements  146 ′ arranged in parallel. Switches  202  are individually controlled by a control line  152  and allow the elements to be selectively couple to ground  204 . 
       FIG. 3  is a graph showing a charge transfer function  302 , denoted as T(t). The charge transfer function  302  corresponds to the proportion of the total charge in a capacitive element that transferred to or discharged from the element as a function of time, t. The charge transfer begins at time t=t 0  and the transfer is substantially discharged at time t=t 1 . In general, the charge transfer is not instantaneous, since the capacitive element, or the connections to it, have a finite resistance. The higher the resistance, the lower the initial slope of the charge transfer function  302 . The time constant, that is the time for the capacitor to charge to 63.2% of its full charge, is equal to the produce of the resistance and the capacitance. Generally, although never 100% charged, the capacitor is considered to be fully charged after five time constants. 
       FIG. 4  is a graphical representation of clock signals and an integrator voltage in a capacitance-to-digital converter, plotted as a function of time. A first clock signal  402  (φ 1 ) controls the excitation signal supplied to the capacitive element to be sensed and a second clock signal  404  (φ 2 ) controls integration of charges. In this example, the second clock signal  404  is the complement or inversion of the first clock signal  402  and both clock signals are symmetrical. The clock signals control transmission gates, as described above. In this example, high signal level indicates that a gate is open and allows transmission (or, equivalently, that a switch is closed), while a low signal level indicates that the gate is closed. The bottom graph  406 , labeled ‘V’, shows an example of a voltage level resulting from integration of a charge. Generally, the voltage level will increase or decrease during the integration period depending on the sign of the charge being integrated. 
     In prior converters, the speed of conversion is limited by the discharge time, t 1 -t 0 . 
     In operation, the average integrated charge over the conversion period is
 
 Q=C   SENS   ×V   REF   −C   OFF   ×V   REF   −R×C   REF   ×V   REF   (1)
 
where V REF  is the reference voltage, C SENS , C REF , C OFF  are the sensor, reference and offset capacitances and R is difference between the number of ones and the number of zeros in the output sequence  102 , divided by the total number of values in the output sequence. This expression assumes that the sampling and integration times are sufficient for the capacitive elements to be fully charged and discharged and that the same reference voltage is applied to each capacitive element.
 
     The integrated charge is controlled, by the binary output sequence, to be zero, so the capacitance of the sensed capacitive element is given by
 
 C   SENS   =C   OFF   +R×C   REF   (2)
 
     Since the ratio R is less than or equal to one, inclusion of the offset capacitance allows sensor capacitances larger than the reference capacitance to be measured. 
       FIG. 5  is graphical representations of a first clock signal  502  and a second clock signal  504  in a capacitance-to-digital converter. In accordance with one embodiment of the invention, a first clock signal  502  has a cycle  506  (denoted as ‘Tcycle’ in the figure). In each cycle, the clock signal has a logic value one for a first interval  508  (denoted as Ts) and logic value zero for a second interval  510  in each cycle of the first clock signal. The first clock signal is used to generate an excitation signal that has a first voltage during the first interval  508  of the first clock signal and a second voltage value (which may correspond to an electrical ground) during the second interval  510  of the first clock. The excitation signal is supplied to the sensed capacitive element to accumulate an electrical charge on the capacitive element. The accumulated charge is dependent upon the applied voltage, so the capacitive element is said to have ‘sampled’ the applied voltage. In each clock cycle  512  of the second clock, the second clock signal  504  has a logic value one for a first interval  514  (denoted as ‘Ti’) and logic value zero for a second interval  516  in each cycle of the second clock signal. A reference electrical charge is generated dependent upon the current binary value of the binary sequence and a combination of the reference charge and the electrical charge on the capacitive element is integrated during the first interval  514  of the second clock to produce an integrated electrical charge. This integrated charge is compared to a threshold to obtain a next binary value of the binary sequence. The binary value is dependent upon whether the integrated electrical charge is above or below the threshold. 
     In one embodiment of the invention, the cycle  506  of the first clock signal has a non-constant duration and the cycle of the second clock signal has a non-constant duration. The varying clock rate reduces electromagnetic radiation from the converter and reduces the sensitivity of the converter to external noise. Such clock signals with varying cycle times are termed ‘spread spectrum clocks’, since the spectrum of the signal is spread across a range of frequencies. 
     In general, when the clock rate is high enough, the capacitances may not have time to fully charge or discharge during a clock cycle. We denote the proportion of maximum charge transferred to the sensed capacitive element during the sample interval as T 1  and denote the proportion transferred from the element during the integration interval as T 2 . Assuming that the charge transfer functions T 1 , T 2  are both constant over the conversion period, the average integrated charge is expressed as
 
 Q=T   2   ×T   1   ×C   SENS   ×V   REF   −C   OFF   ×V   REF   −R×C   REF   ×R   REF   (3)
 
     Where, as before, V REF  is the reference voltage, C SENS , C REF , C OFF  are, respectively, the sensor, reference and offset capacitances, and R is ratio of ones to zeros in the output stream  102 . It is assumed that the offset and reference capacitance circuits have very little resistance, so that the capacitors are fully charged and discharged in each cycle. 
     The integrated charge is controlled, by the binary output sequence, to be zero. Hence, the capacitance of the sensed capacitive element is 
     
       
         
           
             
               
                 
                   
                     
                       C 
                       SENS 
                     
                     = 
                     
                       
                         1 
                         
                           
                             T 
                             1 
                           
                           ⁢ 
                           
                             T 
                             2 
                           
                         
                       
                       × 
                       
                         [ 
                         
                           
                             C 
                             OFF 
                           
                           + 
                           
                             R 
                             × 
                             
                               C 
                               REF 
                             
                           
                         
                         ] 
                       
                     
                   
                   , 
                 
               
               
                 
                   ( 
                   4 
                   ) 
                 
               
             
           
         
       
     
     Since the ratio R is less than or equal to one, inclusion of the offset capacitance allows larger sensor capacitances to the measured. In addition, if the integration times are small enough, the sensitivity of the converter can be adjusted by controlling the ratio of sampling times. 
     In a first embodiment of the invention, the clock rate is varied but the rate is controlled to be sufficiently slow at all times that the proportions T 1 ,T 2  are substantially equal to unity. This condition satisfies the assumption that the charge transfer functions T 1 ,T 2  are constant over the conversion period. It is noted that the proportions T 1 ,T 2  depend upon the electrical properties of the sensed capacitive element and the connections to it. Thus, the minimum cycle duration may be selected dependent upon the electrical properties of the sensed capacitive element and the connections to it. 
       FIG. 6  is a graphical representation of a first clock signal  502 , a second clock signal  504  and an integrator voltage  406  in a capacitance-to-digital converter in accordance with some embodiments of the invention. In  FIG. 6 , the first clock signal  502  and the second clock signal  504  have variable duration clock cycles. However, the minimum sample interval Ts and the minimum integration interval Ti are long enough that the sensed capacitive element has time to substantially charge and discharge in each cycle. As a result, the change in the integrator voltage  406  depends only on the comparator output signal, and is independent of the clock cycle period. 
     In a further embodiment of the invention, the clock rate is varied (i.e. a spread spectrum clock is used), but the integration and sampling times are kept constant over the conversion time. The clock is therefore asymmetric. Again, this satisfies the assumption that the charge transfer functions T 1 , T 2  are constant over the conversion period. 
     The charge transfer functions, T 1  and T 2 , are functions of time and depend upon the electrical characteristics of the sensor. If these characteristics are known (for example from calibration measurements) an absolute value of the sensed capacitance may be obtained. Otherwise, since the scaling factor 
             1       T   1     ⁢     T   2             
is constant, the expression may be used to compare a number of capacitive elements with similar charge transfer characteristics—such as the elements of a capacitive touch screen.
 
       FIG. 7  is a graphical representation of a first clock signal  502 , a second clock signal  504  and an integrator voltage  406  in a capacitance-to-digital converter in accordance with some embodiments of the invention. In  FIG. 7 , the first clock signal  502  and the second clock signal  504  have variable duration clock cycles. However, the sample interval Ts and the integration interval Ti are constant from one cycle to the next during a conversion. As a result, the change in the integrator voltage  406  depends only on the comparator output signal, and is independent of the clock cycle period, even though the sensed capacitive element may not have time to fully charge and discharge (in the figure, the integrated voltage level is still rising or falling at the end of each integration period). 
       FIG. 8  is a simplified block diagram of a touch and proximity sensor in accordance with some embodiments of the invention. The sensor comprises a number of row elements, R 1 , R 2 , R 3 , . . . , R m , and a number of column elements C 1 , C 2 , C 3 , . . . , C n , arranged to form a grid. The elements may be positioned in a touch screen of a portable electronic device, for example. A capacitance-to-digital converter  100  converters a capacitance of the sensor to a binary sequence  102  that may be filtered and decimated in digital filter  110  to provide a multi-bit, digital representation  804  of the capacitance. The digital representation  804  may be input to a processor to provide a man-machine interface, for example. The converter  100  supplies an excitation signal  106  that is amplified in amplifier  806  and fed to column switch  808 . The column switch is controlled to select the column element to which the excitation is applied. Similarly, a row switch  810  is controlled to select the row element that is to be sensed. The sensed signal  108  is passed back to the converter  100 . In operation, each row-column pair is selected in turn. 
       FIG. 9  is further block diagram of a touch and proximity sensor in accordance with some embodiments of the invention.  FIG. 9  shows a more detailed view of a section of touch and proximity sensor. In this example, each row and each column comprise a connected line of diamond shaped conductors. The matrix of sensing elements is sometimes referred to as a Transparent Diamond Matrix and may be stacked on top of or integrated with a display of an electronic device. When a capacitive object, such as a finger, stylus, pen etc. is touching the sensor or is moved into the proximity of the sensor in a region  902 , the capacitance  904  between row and column elements in the region is increased. In the example shown, the capacitance is increased for the four element pairs (C i-1 , R j-1 ), (C i , R j-1 ), (C i-1 , R j ) and (C i , R j ). If the capacitive object is moved away slightly, more elements pairs will be affected, but the change in capacitance for each pair will be reduced. Thus, for sensing both touch and proximity, the converter  100  should be able to measure a wide range of capacitances. The use of an offset capacitance is one way of achieving this. Additionally, since the capacitance changes may be small, the converter should be insensitive to noise. Use of a variable speed clock reduces the converter&#39;s sensitivity to noise. Still further, for a sensor with a high spatial resolution, many element pair must be sensed for each position measurement, so it is advantageous for the converter sensor to have a short conversion time. 
     In one embodiment, the first and second clock signals are generated from a pseudo-random binary sequence or random binary sequence. An example is shown in  FIG. 10 . A random or pseudo-random sequence  1002  is generated. Circuits for generating pseudo-ransom sequences are well known to those of ordinary skill in the art. At each rising edge of the pseudo-random sequence  1002 , a fixed duration pulse is generated in a first clock signal  502 . At each falling edge of the pseudo-random sequence  1002 , a fixed duration pulse is generated in a second clock signal  504 . In this manner, two interleaved, but uncorrelated, clock signals are obtained. These may be used as Φ 1  and Φ 2  (or Φ 2  and Φ 1 ). Other methods for generating the spread spectrum clock signals Φ 1  and Φ 2  will be apparent to those of ordinary skill in the art. 
     In the foregoing specification, specific embodiments of the present invention have been described. However, one of ordinary skill in the art appreciates that various modifications and changes can be made without departing from the scope of the present invention as set forth in the claims below. Accordingly, the specification and figures are to be regarded in an illustrative rather than a restrictive sense, and all such modifications are intended to be included within the scope of the present invention. The benefits, advantages, solutions to problems, and any element(s) that may cause any benefit, advantage, or solution to occur or become more pronounced are not to be construed as a critical, required, or essential features or elements of any or all the claims. The invention is defined solely by the appended claims including any amendments made during the pendency of this application and all equivalents of those claims as issued.