Abstract:
A frequency dithering circuit reduces emissions that cause Electro-Magnetic Interference (EMI) by spreading the spectrum of a clock. The clock sequences a counter that drives a digital count value to a digital-to-analog converter (DAC). The DAC outputs a sawtooth wave with a wide voltage swing. A subtractor scales down the voltage swing to produce a reduced-swing sawtooth wave which is used as an upper limit voltage. Comparators trigger a set-reset latch to toggle the clock when current pumps charge and discharge a capacitor beyond voltage limits. Since the upper limit voltage is the reduced sawtooth wave from the subtractor, the amount of time to charge the capacitor varies, dithering the period of the clock. The degree of dithering can be adjusted by programming the feedback resistance in the subtractor. The subtractor reduces the sensitivity of dithering to errors in the DAC, allowing for an inexpensive, less precise DAC.

Description:
FIELD OF THE INVENTION 
     This invention relates to Electro-Magnetic Interference (EMI) reduction, and more particularly to circuits with frequency dithering for reduced EMI. 
     BACKGROUND OF THE INVENTION 
     Power converters may include a large transistor that is rapidly switched on and off. This rapid switching can cause noise on the resulting power-supply voltage, and upon other nearly signals. Downstream and upstream devices from the power converter can generate emissions that create disturbances in other electronic equipment. For example, a noisy switching power supply driving a portable computer may emit radiation that interferes with a television receiver. Intentional receivers, such as televisions, cellular phones, pagers, and wireless devices, are often affected by unintentional transmitters that emit electromagnetic radiation. As both types of electronic equipment become more common, interference becomes more noticeable to the consumer. 
     Electromagnetic interference (EMI) is a measure of the amount of interference that an electronic device (the unintentional transmitter) disturbs an intentional receiver. Government agencies such as the Federal Communication Commission (FCC) strictly regulate the amount of radiation or EMI that an electronic device can generate. 
     Improving technology also worsens the EMI problem. Faster clock rates of higher-speed equipment generate more radiation. Higher resolution monitors and displays require that more pixels be transferred to the screen for each screen refresh period; thus a higher clock rate and more interference results. 
     Traditional techniques to reduce EMI attempt to contain radiation or to reduce the amount of radiation generated. Coax wires and shielded cables are effective at containing radiation, but are expensive, heavy, bulky, and inflexible. The weight and bulk of shielded cables make them undesirable for portable devices. Metal chassis with sealed seams are effective for reducing EMI of desktop equipment, but portable devices are kept light by using plastic. 
     Lower voltages reduce the intensity of the radiation generated, and the new 3-volt standard has helped reduce EMI at all harmonics. Proper impedance matching and termination of signals reduces ringing and harmonics, and shorter signal traces further reduce radiation. Ground planes on PCB&#39;s or ground lines running parallel with signal lines effectively shield signals on boards. Filtering can reduce sharp rise and fall times and reduce radiation by wave shaping since more sinusoidal waves have fewer harmonics than square waves. Of course, filters require additional capacitors, resistors, or inductors, raising the cost. All of these techniques are useful to varying extents. 
     Large physical components are often needed as filters to reduce noise and EMI. For example, a large inductor coil may be added to a power converter, along with high-value capacitors that also are bulky. Precision capacitors or resistors may be needed. These bulky components are undesirable and costly and defeat integration. 
     A newer technique to reduce EMI is to vary or modulate the frequency of clocks. This technique known as spread spectrum, since the frequency spectrum of the clock is spread out over a wider range of frequencies.  FIG. 1  shows a graph of radiation intensity as a function of frequency for an un-modulated clock signal. A sharp spike occurs at a harmonic of the clock&#39;s frequency, 40 MHz. Since the clock constantly operates at the rated frequency, all of the energy of the radiation appears in a narrow spike, which has a large amplitude. The spike has an amplitude over the EMI limit set by the FCC. The high intensity of the spike can cause interference in a receiver. 
       FIG. 2  is a graph of radiation intensity as a function of frequency generated by a modulated clock. The clock&#39;s frequency is not constant, but is varied with time over a range of +5% to −5% of the rated frequency. Thus the clock operates at 40 MHz for a period of time, but also operates at other frequencies between 38 MHz and at 42 MHz at other times. Such a clock can be generated by slowly changing the frequency from 38 MHz to 42 MHz and then slowly reducing the frequency back to 38 MHz. A voltage-controlled oscillator (VCO) can be used with the input voltage being slewed back and forth between voltages that generate 38 MHz and 42 MHz oscillations. 
     Since the modulated clock spends only part of the time at 40 MHz, the intensity of the radiation, averaged over a relatively long time, is reduced. The total energy of the radiation at all frequencies is about the same as for the un-modulated clock of  FIG. 1 , but the intensity at any particular frequency is greatly reduced. Interference at any one frequency is reduced since receivers generally are tuned to a particular frequency (even FM receivers are tuned to a relatively small range of frequencies). 
     Thus modulating the clock&#39;s frequency reduces the maximum intensity of radiation at any one frequency, although the energy radiated at all frequencies is not reduced. This has the practical effect of reducing interference for receivers tuned to a fixed frequency. 
     Long Sweep Period of Modulation— FIG. 3   
       FIG. 3  is a graph of a modulated clock&#39;s frequency as a function of time over a few sweep periods. The clock&#39;s nominal frequency is 40 MHz. The clock is modulated by +/−5%, from 38 MHz to 42 MHz. The clock&#39;s frequency is swept from minimum to maximum frequencies over one or two thousand clock periods so that adjacent clock pulses have a very small variation. A 40 MHz clock with a 25 nanosecond (ns) period is varied from 26.25 ns to 23.75 ns over a sweep period, a variation of +/−1.25 ns. A 37 KHz sweep rate has a sweep period of 27 micro-seconds (μs). A sweep period is 27 μs/25 ns or 1081 clock periods. The cycle-to-cycle period variation for two adjacent clock periods is thus 5 ns/1081 or 4.62 pico-seconds (ps). The sweep frequency is typically 15 to 50 KHz. 
     Such spread spectrum using frequency dithering is effective in reducing EMI. Bulky filter components such as large capacitors and inductors are not needed. However, often such frequency dithering requires switching precise components such as precision capacitors to generate small steps or adjustments in frequency. Matching of these precision capacitors or resistors is difficult. Very large capacitors or very small currents or resistors must be used. Such small currents are themselves subject to noise interference and leakages and are thus undesirable. 
     What is desired is a frequency dithering circuit for reducing EMI for a switched power supply. A dithering circuit that does not need bulky filters or precision components is desirable. A dithering circuit that can be integrated with other circuits for a small form factor is desirable. A dithering circuit that can be tuned or programmed for the degree of frequency dithering is desirable. A dithering circuit that can be used with an oscillator for a switched power supply is desirable. A dithering circuit that can be used as a module for a clock generator is also desired. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  shows a graph of radiation intensity as a function of frequency for an un-modulated clock signal. 
         FIG. 2  is a graph of radiation intensity as a function of frequency generated by a modulated clock. 
         FIG. 3  is a graph of a modulated clock&#39;s frequency as a function of time over a few sweep periods. 
         FIG. 4  is a schematic of a frequency dithering circuit for modulating oscillating of a clock. 
         FIG. 5  is a diagram of the control circuit. 
         FIG. 6  shows a simple DAC. 
         FIG. 7  is a waveform graph of sawtooth voltages for controlling frequency dithering. 
         FIG. 8  is an expansion of the time scale of the waveform of  FIG. 7 . 
         FIG. 9  is a spectrum diagram of an oscillator without frequency dithering. 
         FIG. 10  is a spectrum diagram of a clock with frequency dithering, such as the circuit of  FIG. 4 . 
         FIG. 11  is a programmable dithering circuit. 
         FIGS. 12A-C  are waveforms showing the effects on emissions of different programmable values of feedback resistance. 
     
    
    
     DETAILED DESCRIPTION 
     The present invention relates to an improvement in EMI reduction circuits. The following description is presented to enable one of ordinary skill in the art to make and use the invention as provided in the context of a particular application and its requirements. Various modifications to the preferred embodiment will be apparent to those with skill in the art, and the general principles defined herein may be applied to other embodiments. Therefore, the present invention is not intended to be limited to the particular embodiments shown and described, but is to be accorded the widest scope consistent with the principles and novel features herein disclosed. 
     The inventors have realized that a digital-to-analog converter (DAC) can be used to adjust the frequency of an oscillator or other clock generator. This frequency dithering can reduce or eliminate bulky filter components and allow for integration and small form factor devices. 
     The inventors further realize that matching of precision resistors and capacitors for the DAC can be relaxed if the resolution requirement of the DAC is relaxed. The DAC resolution can be reduced by scaling and subtraction of the DAC output. The inventors further realize that programming the subtractor after the DAC can allow for the degree of dithering to be programmed. 
       FIG. 4  is a schematic of a frequency dithering circuit for modulating oscillating of a clock. Clock CLK may be used to switch a large field-effect transistor (FET) in a switching power supply, or may be used for some other purpose. SR latch  24  toggles to cause its outputs CLK and CLKB to pulse. The CLK signal is fed back to control circuit  22 . In some embodiments, control circuit  22  is a counter, such as a binary counter. The count value from control circuit  22  is input to digital-to-analog converter (DAC)  20 , and DAC  20  converts the digital count value to an analog voltage VDAC that is a fraction of analog input voltage Vref, with the fraction depending on the digital input value from control circuit  22 . 
     Voltage divider  26  generates a lower voltage Vref 2  from Vref, and this lower reference voltage Vref 2  is input to subtractor  31 . The other input to subtractor  31  is the converted analog voltage VDAC from DAC  20 . 
     Subtractor  31  includes op amp  30 , which receives Vref 2  on its non-inverting (+) input. Input resistor  36  is in series between VDAC from DAC  20  and the inverting (−) input to op amp  30 . Feedback resistor  18  is connected between the output and inverting input of op amp  30 . The ratio of resistances of input resistor  36  and feedback resistor  18  determines the amount of scaling of VDAC, which also determines the degree of frequency dithering. 
     Subtractor  31  scales down the analog output from DAC  20 . The output of subtractor  31  is ramp voltage Vrmp_max, which is a scaled-down reproduction of the VDAC voltage from DAC  20 . Scaling down the DAC output also reduces the effect of errors from DAC  20  on the frequency of clock CLK. Thus the dithering circuit is less sensitive to errors from DAC  20  due to subtractor  31 . A larger step size or difference between voltage levels of DAC  20  can be tolerated. Thus resistors, capacitors, or other components inside DAC  20  do not have to be matched as precisely. 
     The binary count value from control circuit  22  counts up and then counts down. The resulting converted analog voltage VDAC from DAC  20  is a sawtooth wave that slowly rises and then slowly falls. Subtractor  31  reduces the amplitude of the sawtooth wave from DAC  20 , so that ramp voltage Vrmp_max is also a sawtooth wave, but with a smaller voltage swing. 
     When CLK is high and CLKB low, switch  46  is closed, and switch  47  is open. Switch  46  allows current source  42  to drive current from the power supply to node Vrmp, charging capacitor  48 . Voltage Vrmp rises. Once Vrmp rises above ramp voltage Vrmp_max, comparator  32  is triggered, since its + input is Vrmp and its − input is Vrmp_max. Comparator  32  triggers a pulse to the reset input of SR latch  24  causing CLK to go low and CLKB to go high. 
     When CLK is low and CLKB is high, switch  46  is open, and switch  47  is closed. Switch  47  allows current sink  44  to sink current to ground from node Vrmp, discharging capacitor  48 . Voltage Vrmp falls. Limit voltage VP is generated by voltage source  28 . Once Vrmp falls below limit voltage VP, comparator  34  is triggered, since its + input is VP and its − input is Vrmp. Comparator  34  triggers a pulse to the set input of SR latch  24  causing CLK to go high and CLKB to go low. 
     Ramp voltage Vrmp_max acts as a maximum voltage limit, while limit voltage VP acts as a minimum voltage limit. While the minimum voltage limit is fixed, the maximum voltage limit varies with the sawtooth wave from DAC  20 . 
     As Vrmp slowly rises as the count value input to DAC  20  increases, the upper limit rises. Since current source  42  delivers a fixed current, it takes a longer period of time to charge capacitor  48  up to the new higher limit of Vrmp. This longer time required to charge capacitor  48  to the higher Vrmp_max delays the next falling transition of CLK, thus increasing the clock period and decreasing the frequency. Likewise, the increased charge on capacitor  48  at the higher Vrmp that matches Vrmp_max takes a longer time to discharge, so the rising transitions of CLK are also delayed. Thus both clock edges are delayed as Vrmp_max rises, and frequency drops. 
     Once control circuit  22  has reached the maximum count value, it reverses direction and counts down. The decreasing digital values applied to DAC  20  produce a series of decreasing steps down in voltage VDAC. These large steps in VDAC are scaled down by subtractor  31  to smaller steps downward in voltage Vrmp_max. As Vrmp_max drops, charging and discharging require less time, and the clock frequency increases. 
     The peak value of Vrmp_max, Vrmp_peak, is related to Vref 2 , VDAC, the series resistance Rs of input resistor  36  and the feedback resistance Rf of feedback resistor  18  as follows:
 
 V rmp_peak= V ref2−(Rf/Rs)*VDAC.
 
     Thus adjusting the ratio of resistors  18 ,  36  adjusts the peak limit voltage, the maximum charging time, and thus the period of the clock. Thus Rf/Rs adjusts a degree or maximum amount of frequency dithering. 
       FIG. 5  is a diagram of the control circuit. Control circuit  22  of  FIG. 4  is a binary up/down counter in some embodiments. An (N+1)-bit binary count value CB 0 :CBN is generated from input clock CLK, CLKB. The inverse count value is CB 0 B:CBNB. This count value CB 0 :CBN is input to DAC  20 . 
     An intermediate count value B 0 :BN and its inverse bits B 0 B:BNB are generated by flip-flops  62 . CLK is applied to the first of flip-flops  62 , and each flip-flop  62  acts as a toggle flip-flop with its QB output fed back to its D input. The Q output of one flip-flop  62  is input as the clock to the next flip-flop  62 . Flip-flops  62  act as a ripple counter producing a binary count value. 
     A final count value is generated from the intermediate count value by logic including AND gates  64 ,  66 , OR gates  68 , and flip-flops  70 . AND gates  64  and  66  act as gate logic to pass either signal B 0 :B(N−1) or B 0 B:B(N−1)B to the D input of D-flip-flops  70  depending the logic signal of BN and BNB. When CLKB is rising from 0 to 1, flip-flop  70  clocks the input signal at D to its output Q and QB. The result is that the count value counts up in binary to FF, then counts down from FF to 00. 
       FIG. 6  shows a simple DAC. Analog input voltage Vref is applied to a voltage divider of resistors  72  to ground, producing a series of voltages between Vref and ground. When resistors  72  all have about the same value, a series of intermediate voltages are produced. Mux logic  80  selects one of these intermediate voltages for output as VDAC. The select for mux logic  80  is the binary count value CB 0 :CBN and its inverse CB 0 B:CBNB from control logic  22 . 
     DAC  20  can be a simplified DAC since subtractor  31  reduces the value of the voltage steps, so that errors produce a smaller effect on the frequency of CLK. Thus frequency dithering is more tolerant of errors. Resistors  72  do not have to be precisely matched, since some error is tolerable. Smaller currents may be used, reducing power. For example, resistors  72  may be matched in resistance values to within 5% of each other when errors are tolerated. 
       FIG. 7  is a waveform graph of sawtooth voltages for controlling frequency dithering. VDAC is a sawtooth wave that slowly swings between ground and 2.5 volts. Subtractor  31  scales this down to a swing of only about 300 mV, from 1.1 to 1.4 volts on Vrmp_max. 
       FIG. 8  is an expansion of the time scale of the waveform of  FIG. 7 . VDAC slowly steps down each time clock CLK pulses, as the count value from control circuit  22  is reduced. Vrmp_max slows steps upward, but by smaller increments due to the attenuation by subtractor  31 . The reverse behavior occurs for the other half of the sawtooth wave, when VDAC is stepping upward. and Vrmp_max is stepping downward. 
       FIG. 9  is a spectrum diagram of an oscillator without frequency dithering. In this idealized example, noise and emissions vary with frequency. The maximum emission occurs at about 80 kHz. The maximum emission is greater than 0 dBV. This may exceed emission limits. 
       FIG. 10  is a spectrum diagram of a clock with frequency dithering, such as the circuit of  FIG. 4 . In this example, noise and emissions vary with frequency. The maximum emissions occurs in a broad range of about 75-90 kHz. The maximum emission is about 15 dBV less than 0 dBV. This may meet emission limits. Thus frequency dithering spread out the peak emission over a wider range of frequencies, producing a broader and lower peak. 
       FIG. 11  is a programmable dithering circuit. The circuit is similar to that of  FIG. 4 , except that feedback resistor  18  is replaced by digitally-controlled resistor  50 . The value Rf of the feedback resistance is controlled by digital select value SEL, which selects one or more parallel resistors using switches. Thus the value of Rf may be programmed. 
       FIGS. 12A-C  are waveforms showing the effects on emissions of different programmable values of feedback resistance. Using digitally-controlled resistor  50  ( FIG. 11 ), the ratio of feedback resistor  18  to input resistor  36 , Rf/Rs, can be adjusted. This varies the contribution of feedback. The amount of frequency dithering is also varied as Vrmp_max swings over a wider voltage range for higher values of Rf/Rs. 
     In  FIG. 12A , Rf/Rs is 0.03. A narrow peak emission of about −15 dBV occurs around 77 kHz. In  FIG. 12B , Rf/Rs is 0.06. The peak emission is lowered to about −20 dBV but is spread over a wider range of frequencies. 
     In  FIG. 12C , Rf/Rs is 0.09. The peak emission is lowered to less than −20 dBV but over a wider range of frequency. 
     Alternate Embodiments 
     Several other embodiments are contemplated by the inventors. For example, while the use of the dithering circuit for a power supply has been described, the dithering circuit may also be used in other power converters, or in more general applications such as a clock generator module or other clock generators. The charge pumps may be implemented as transistors as approximations. 
     A square wave, a sine wave, or other kinds of waveforms for modulation can be generated instead of the sawtooth wave described. The entire circuit may be integrated onto a single integrated circuit. Polarities of signals can be reversed. Other counting and encoding schemes other than binary may be used. The control logic can be implemented as a binary counter using synchronous rather than ripple logic, and can be implemented in a variety of ways. The DAC can be implemented in a variety of ways. Voltages and other data shown in the waveforms are examples and can vary with different embodiments and conditions. Reset inputs (not shown) can be added to each flip-flop  62 ,  70  in  FIG. 5 . Other kinds of bistables may be substituted for the SR latch, such as JK flip-flops, toggle flip-flops, D-type flip-flops, etc. 
     The background of the invention section may contain background information about the problem or environment of the invention rather than describe prior art by others. Thus inclusion of material in the background section is not an admission of prior art by the Applicant. 
     Any methods or processes described herein are machine-implemented or computer-implemented and are intended to be performed by machine, computer, or other device and are not intended to be performed solely by humans without such machine assistance. Tangible results generated may include reports or other machine-generated displays on display devices such as computer monitors, projection devices, audio-generating devices, and related media devices, and may include hardcopy printouts that are also machine-generated. Computer control of other machines is another tangible result. 
     Any advantages and benefits described may not apply to all embodiments of the invention. When the word “means” is recited in a claim element, Applicant intends for the claim element to fall under 35 USC Sect. 112, paragraph 6. Often a label of one or more words precedes the word “means”. The word or words preceding the word “means” is a label intended to ease referencing of claim elements and is not intended to convey a structural limitation. Such means-plus-function claims are intended to cover not only the structures described herein for performing the function and their structural equivalents, but also equivalent structures. For example, although a nail and a screw have different structures, they are equivalent structures since they both perform the function of fastening. Claims that do not use the word “means” are not intended to fall under 35 USC Sect. 112, paragraph 6. Signals are typically electronic signals, but may be optical signals such as can be carried over a fiber optic line. 
     The foregoing description of the embodiments of the invention has been presented for the purposes of illustration and description. It is not intended to be exhaustive or to limit the invention to the precise form disclosed. Many modifications and variations are possible in light of the above teaching. It is intended that the scope of the invention be limited not by this detailed description, but rather by the claims appended hereto.