Abstract:
A high performance digitalized Programmable Gain Amplifier (PGA). In prior art circuit, a dual-ladder DAC is employed for gain control, the back gate leakage of NMOS resistors in the fine ladder conquers fine ladder nominal current and it produces non-monotonic gain scallop. Two new art design techniques: (1) adaptively control the fine ladder; and (2) use dummy PMOS brunch device leakage compensates for the NMOS resistor device leakage, are proposed so that the non-monotonic scallops are substantially eliminated and 13-bit resolution/accuracy PGA has been achieved.

Description:
PRIORITY 
     This application claims priority to U.S. Provisional Application No. 61/595,340, filed Feb. 6, 2012, entitled “BEMF Monitor Gain Calibration Stage in HDD Servo IC”, which is incorporated by reference in its entirety. This application also claims priority to U.S. Provisional Application No. 61/625,485, filed Apr. 17, 2012, also entitled “BEMF Monitor Gain Calibration Stage in HDD Servo IC”, which is also incorporated by reference in its entirety. 
    
    
     TECHNICAL FIELD 
     This application is directed, in general, to amplifiers in Hard Disk Drives (HDD) servo system and, more specifically, to a gain calibration of an amplifier in monitoring back electro-motive force (BEMF) of VCM motor in servo integrated circuits (ICs). 
     BACKGROUND 
       FIG. 1  illustrates a prior art BEMF system  100  having an A 1  gain amplifier  110  of a Voice Coil Motor (VCM)  120  BEMF monitor in HDD servo IC. 
     In gain calibration mode, the VCM motor  120  is made still so its BEMF voltage is zero. Meanwhile, about 100 mA current is forced flowing through VCM motor and sense resistor Rs. The voltage drop on the sense resistor Rs is amplified by a digital programmable gain amplifier A 1 , and the A 1  gain output is cancelled out with VCM voltage at A 2  gain stage  130 . Make A 2  output zero by digitally changing A 1  gain, A 1  amplifier  110  gain is digitally calibrated to be equal to a ratio of motor resistance R VCM  and sense resistance R SNS , that is, 
     
       
         
           
             
               A 
               ⁢ 
               
                   
               
               ⁢ 
               1 
               ⁢ 
               Gain 
             
             = 
             
               
                 
                   R 
                   VCM 
                 
                 
                   R 
                   SNS 
                 
               
               . 
             
           
         
       
     
     In BEMF normal operation mode while VCM BEMF voltage presents, the current-resistance-product item of Rs is gained up by the calibrated A 1  gain and is subtracted from the voltage across the VCM at A 2  stage, that is,
 
 V   A2OUT   =V   VCM   −V   A1OUT =( V   VCM     —     BEMF   +I×R   VCM )− I   VCM   ×R   SNS   ×A 1Gain= V   VCM     —     BEMF   Equation (1)
 
As shown in Equation (1), A 2  amplifier  130  output is an estimation of VCM motor BEMF voltage. In modern HDD servo IC circuits, a 13-bit DAC is used for A 1  programmable gain stage  110  to achieve required mV-level BEMF voltage measurement accuracy.
 
       FIGS. 2A and 2B  illustrate two prior art Programmable Gain Amplifier (PGA) approaches to BEMF monitor gain calibration of A 1  gain  110 . In  FIG. 2A , switches of a first prior art PGA  200  performing gain programmability are tied to an operational amplifier&#39;s (OPA&#39;s) negative input node, and the switch on-state resistance R ds,on  does not contribute to PGA gain error, because of an infinite impedance, thus zero input bias current to the inputs  211  and  212  of CMOS OPA. 
     However, the scheme of the PGA  200  also has a disadvantage. The gain is not linear to resistance increment/decrement which is linearly coded by switches S 0 -S(N). For example, in  FIG. 2A , suppose N=63 (i.e., 64 switches/resistors in total) and the resistance between every node tapped out by switches are evenly equal to a given resistance R. When the digital input changes from code X=‘00H’ (switch S 0  is turned on and all others are off) to X=‘01H’ (switch S 1  is turned on and all others are off) and then to X=‘02H’ (switch S 2  is turned on and all others are off), the PGA  200  gain, which is R f /R in , moves from 63 to 62/2 then to 61/3 which are non-linear gain steps. In other words, the PGA  200  does not have a linear transfer function although the digital code linearly changes. This lack of linearity does not meet system level requirements for typical servo systems. 
     In  FIG. 2B , an alternative prior art PGA  250  of the A 1  gain  110 , the input resistance R in  between input Vin and the OPA negative input node is fixed, and the feedback resistance R f  is linearly coded by switches. As a result, the gain of the PGA  250  of an A 1  PGA  110  is a linear function of the linearly coded resistance. For example, when digital code X=“00H” (switch S 1  is turned on and all other switches are off) moves to X=“01H” (S 2  is turned on and all others are off) then to X=“02H” (switch S 3  is turned and all others are off) and finally moves to X=“3FH” (switch S( 64 ) is turned on and all others are off), the gain R f /R in  changes from to 
               R   +     R       S   ⁢           ⁢   1     ,   on           R   in           
to
 
                   2   ⁢   R     +     R       S   ⁢           ⁢   2     ,   on           R   in       ⁢           ⁢   then   ⁢           ⁢   to   ⁢           ⁢         3   ⁢   R     +     R       S   ⁢           ⁢   3     ,   on           R   in       ⁢           ⁢   and   ⁢           ⁢   finally   ⁢           ⁢           64   ⁢   R     +     R       S   ⁢           ⁢   64     ,   on           R   in       .           
The A 1  gain is linear to the digital code change only when the switch on-state resistances are ideally zero.
 
     Disadvantageously, however, in the prior art PGA  250  of  FIG. 2B , the switch on-state resistances S 1 -SN, are in the signal path and they contribute to the amplifier gain error, unlike the prior art PGA  200 . Furthermore, the switch resistances S( 1 )-S(N) and the poly resistors R(N) that create the gain for the PGA  250  have different temperature coefficients and voltage coefficients, and it can traverse into gain error over temperature variation and over voltage signal excursion during the operation. 
     Moreover, to succeed in high-volume product business such as HDDs, it is important to decrease circuit element counts and silicon area, for such reasons as saving cost of the IC controlling the HDD servo mechanism. In the discussed prior art PGAs, implementing single-ladder architecture of 13-bit programmable gain needs 2 13 =8192 count of resistors and switches. This is unacceptable for most high-volume HDD servo IC applications. 
       FIG. 3  shows a prior art dual ladder DAC, with NMOS resistors used in the fine ladder forming a 13 bit programmable PGA.  FIG. 3  illustrates a prior art dual-ladder architecture of 6+7, that is, implementing 6-bit in a coarse ladder and 7-bit in a fine ladder. This avoids the large number of switches and resistors used in either the PGA  200  or the PGA  250 . The circuit complexity is reduced to 2 6 +2 7 =192 count of switches and resistors, which is about 43 times less complex in terms of used switches/resistors, compared to the single-ladder architecture of 13 bit resolution. 
     In the dual-ladder DAC architecture  300  of the A 1  gain  110 , to minimize DNL/INL error over process and temperature variations, (1) The resistance of on-state switch  321  and  322  between a coarse ladder  310  and a fine ladder  320  are considered as one unit of the fine ladder  320 ; (2) MOS resistors  323 ,  324 ,  325 ,  326  are used in the fine ladder  320 , where a MOS resistor is defined as a MOS device working in deep triode region. Therefore, all the units along the fine ladder are same type of elements and they match with each other very well (over temperature and process variations) and the DAC DNL/INL performance is improved and silicon area is reduced. 
     However, a trade-off exists regards PGA  300  performance and the ratio between the fine ladder  320  total resistance N×R FL  (where N is fine ladder  320  MOS resistor count) and the coarse ladder  310  unit resistance R CL , which is chosen to shunt with the fine ladder  320  for a specific input digital code. If ratio (N×R FL )/R CL  is smaller than a certain value, a “shunt effect” degrades the DAC DNL/INL performance. This is because when the fine ladder  320  is switched and shunted to a specific coarse ladder resistor R 1 , the effective resistance of the shunted unit is relatively less than it used to be. In implementations, the error caused by the shunt effect 
             1   -         R   CL     //     (     NR   FL     )         R   CL             
should be less than 0.5LSB. To reach the goal, R FL  should be designed larger than a specific value. On the other hand, if ratio (N×R FL )/R CL  is too large and over a certain threshold, the nominal current flowing through the fine ladder  320  is so small that it is conquered by the current leakage in the fine ladder  320 .
 
     In PGA  300 , if equal resistances are used in the coarse ladder  310 , the resulting programmable gain is not linear to the digital code as illustrated as line  330 . To obtain the linear gain as line  331 , the resistances in the coarse ladder  310  have to be carefully computed and they must not be equal. 
     As shown in  FIG. 4  of a prior art PGA  400  and as understood and discovered by the present inventors, the fine ladder  420  nominal current is mostly leaked away locally at the back gate parasitic p-n junction diodes  424  and  425  in MOS resistors  421 , back gate parasitic p-n junction diodes  426  and  427  in MOS resistor  422  and so on, as well as back gate parasitic p-n junction of switches  431 ,  432  and so on, all associated with the fine ladder  420 . When ratio (N×R FL )/R CL  is too large and the nominal current flowing through fine ladder  420  is small and comparable to leakage current, the net current goes through fine ladder MOS resistor channel notably decreases due to local MOS back gate leakage. That is, Ids 1 &gt;Ids 2 &gt;Ids 3 &gt;Ids 4  and so on until to some point, the MOS channel current is completely conquered by local leakage current and become zero. After that, the local leakage current is provided by a current coming from a reversed direction. Thus the net current direction flowing through the MOS resistor channel on the right end of the fine ladder  420 , for example devices  426  and  427 , are reversed to what it is supposed to be. The “leakage effect” causes the net current flowing through the prior art fine ladder  420  is not maintained in a consistent direction, therefore the voltage potential along the fine ladder does not monotonically decreases and it produces unacceptable non-monotonic gain scallop. 
     As understood by the present inventors, to achieve 13 bit resolution/linearity and minimize silicon area consumption, which is important to high volume HDD servo products, the error due to ‘shunt effect’ needs to be less than 0.5LSB, thus MOS resistors are used in the DAC fine ladder  420  and its resistances are designed to be much larger than the coarse ladder  410  unit resistance, that is, a large ratio (N×R FL )/R CL . An undesired side-effect, however, is that the nominal current of the fine ladder  420 , which is a ratio R CL /(N×R FL +R CL ) of the total current from Vin to Vout of the PGA stage, become extremely small. It is comparable to or even conquered by the back gate current leakage of the fine ladder  420  MOS resistor  421 ,  422  . . . and DAC switches  431 ,  432  and so on. It causes non-linear scallop and damages 13 bit resolution/linearity performance, especially when the voltage input to the A 1  gain stage is small and therefore the nominal current flowing through fine ladder is very small. 
     Therefore, there is a need in the art as understood by the present inventors to address a design challenge is to balance between the “shunt effect” and the “leakage effect” and remove the non-monotonic scallop, wherein resistance in the coarse ladder is unevenly set to achieve linear coded programmable gain. 
     SUMMARY 
     A first aspect provides an apparatus including a hard drive preamplifier, the hard drive preamplifier including: a coarse ladder having a plurality of resistors, wherein each of the plurality of resistors are individually addressable; a fine ladder that is coupled to the coarse ladder, wherein the find ladder includes a plurality of banks of FETs, each bank including a plurality of FETs, and a fine ladder controller, wherein the each of the plurality of banks of the fine ladder are controllable by the fine ladder controller. 
     A second aspect provides a system including a hard drive preamplifier, the hard drive preamplifier including: a first amplifier having a first gain, the first amplifier including a coarse ladder having a plurality of resistors, wherein each of the plurality of resistors are individually addressable; a fine ladder that is coupled to the coarse ladder, wherein the find ladder includes a plurality of banks of FETs, each bank including a plurality of FETs, and a fine ladder controller, wherein the each of the plurality of banks of the fine ladder are controllable by the fine ladder controller, a second amplifier having a second gain, a summer coupled to an output of the first gain and the second gain. 
     A third aspect provides a system a system including a hard drive preamplifier, the hard drive preamplifier including: a first amplifier having a first gain, the first amplifier including a coarse ladder having a plurality of resistors, wherein each of the plurality of resistors are individually addressable; a fine ladder that is coupled to the coarse ladder, wherein the find ladder includes a plurality of banks of FETs, each bank including a plurality of FETs, and a fine ladder controller, wherein the each of the plurality of banks of the fine ladder are controllable by the fine ladder controller, a second amplifier having a second gain, a summer coupled to an output of the first gain and the second gain, wherein, each of the FETs is an NFET, wherein a PMOS is coupled to each corresponding NFET. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Reference is now made to the following descriptions: 
         FIG. 1  illustrates BEMF monitor  100  with a BEMF digital programmable gain amplifier; 
         FIG. 2A  illustrates a first prior art PGA of the BEMF programmable gain amplifier of  FIG. 1 ; 
         FIG. 2B  illustrates a second prior art PGA of the BEMF programmable gain amplifier of  FIG. 1 ; 
         FIG. 3  illustrates a prior art PGA  300  where design trade off of “shunt effect” and ‘leakage effect” simultaneously exist. 
         FIG. 4  illustrates a prior art PGA  400  where the back gate current leakage conquer the nominal fine ladder current, the voltage potential along the fine ladder does not monotonically decreases and non-linear gain scallop occurs. 
         FIG. 5  illustrates a new art BEMF programmable gain amplifier  500  having a plurality of levels of fine ladder adaptive control and back gate leakage current compensation; 
         FIG. 6  illustrates that the back gate leakage current of NMOS devices associated with node  599  is locally compensated by PMOS devices back gate leakage current. 
         FIG. 7  illustrates a comparison between the prior art gain plot (red trace with non-linear scallop) and the new art gain plot (purple straight line) while digital gain control code is programmed from 0000H to 1FFFH. 
     
    
    
     DETAILED DESCRIPTION 
     Turning to  FIG. 4 , illustrated is one aspect, to avoid the on-state switch resistance effect and achieve 13-bit resolution and accuracy on the programmable gain amplifier, the schematic in  FIG. 2A  (with no Rdson effect on gain accuracy) has been chosen and it is improved to be linear with respect to the programming code. To achieve linear gain in the schematic  FIG. 4 , every resistance unit in the coarse ladder should be carefully calculated. Given a minimum gain G MIN  and a maximum gain G MAX  and a targeted coarse ladder resolution implementation (e.g. 6 bit), following equations (2), (3), (4) should be used to determine every resistor value in the coarse ladder and the result turns out to be R head =5.188 Kohm, R 1 =65.18 ohm, R 2 =66.84 ohm, R 3 =68.56 ohm . . . R 64 =1031 ohm, R tail =13.294 Kohm for a specific design. 
     
       
         
           
             
               
                 
                   
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     Although the coarse ladder resistance calculation in previous paragraph makes the PGA gain linear to the input programming code, there are practical issues with the circuit operation. It is difficult to use a fixed fine ladder design to simultaneously meet PGA accuracy requirement at both following cases. Case (1): when the fine ladder is shunted with R 1  (the minimum resistance of the fine ladder, then the fine ladder current is smallest and easily conquered by leakage, thus the leakage effect dominantly degrades PGA performance and scallop appears), and case (2): when the fine ladder is shunted with R( 64 ) (the maximum resistance of the coarse ladder, then the shunt effect get to its peak point and dominantly degrades the PGA performance). 
     In  FIG. 5 , illustrated is a circuit  500  for new art PGA in servo BEMF monitor circuit. In the circuit  500 , an adaptive control on the fine ladder depends on which coarse ladder resistor is shunted so the ratio (N×R FL )/R CL  is adaptively adjusted, therefore the net current flowing through every unit in the fine ladder is kept relatively constant. 
     For example, when 6 bit MSB decoder process digital input MSB bit &lt; 11 : 6 &gt; and its output turn on switches SWC( 0 )  542  and SWC( 1 )  543 , fine ladder is shunted with the coarse resistor R 1 =65.3 ohm, which is the minimum in the coarse resistance from R( 1 ) to R( 64 ). Meanwhile, the fine ladder adaptive control logic circuit  510  set all output control signal  511 ,  512 ,  513   514  ‘high’, so that all four NMOS resistor banks (respectively with MOS resistor of aspect ratio 8(W/L), 4(W/L), 2(W/L), 1 (W/L), 1 (W/L)) are turned on, and equivalent fine ladder resistance of 64*R 2 /16=4R 2  is shunted with R 1 . Therefore, when the smallest coarse ladder resistance R 1  is chosen to be shunted, the fine ladder resistance is adaptively adjusted be its smallest, to keep the fine ladder current, which is a ratio R 1 /(R 1 +4R 2 ) of total current, be a constant and far away from the level easily conquered by leakage current. 
     When 6 bit MSB decoder process input MSB bit&lt; 11 : 6 &gt; and output turn on switches SWC( 63 )  535  and SWC( 64 )  536 , the fine ladder is shunted with the coarse resistor R( 64 )=1031 ohm, which is the maximum in the coarse resistance from R( 1 ) to R( 64 ). Meanwhile, the fine ladder adaptive control logic circuit  510  set output control signal  511  ‘high’ and other control signal  512 ,  513   514  ‘low’, so that only one fine ladder bank with MOS resistor aspect ratio 1*(W/L) is turned on and all other three NMOS resistor banks (respectively with MOS resistor of aspect ratio 8(W/L), 4(W/L), 2(W/L) are turned off, and equivalent fine ladder resistance of 64*R 2  is shunted with R( 64 ). Therefore, when the largest coarse ladder resistance R( 64 ) is chosen to be shunted, the fine ladder resistance is also adaptively adjusted be its largest, to keep the fine ladder current, which is a ratio R( 64 )/(R( 64 )+64*R 2 ) of total current, be a constant and far away from the level easily conquered by leakage current. 
     In a general case, when 6 bit MSB decoder process input MSB bit&lt; 11 : 6 &gt; and output turn on two switches SWC(N−1) and SWC(N) associated with an arbitrary resistor R(N), the fine ladder is shunted with the coarse resistor R(N). Meanwhile, the fine ladder adaptive control logic circuit  510  set some of control signals  511 ,  512 , 513 , 514  ‘high’ and the rest of control signals ‘low’, so that a combination of fine ladder banks with MOS resistor aspect ratio i*(W/L), where i=1, 2, 3, 4, is turned on and all other NMOS resistor banks are turned off, and an equivalent fine ladder resistance of 64*R 2 /j, where j=1 to 16, is shunted with R(N). Therefore, the fine ladder current, which is a ratio R(N)/(R(N)+64*R 2 /j) of total current, is relatively a constant and far away from the level easily conquered by leakage current. Because the aspect ratio of the fine ladder MOS resistor are binary scaled so an effective of fine ladder resistance of 64*R 2 /j, where j=1, 2, 3, 4 . . . 16 can be adaptively shunted with the corresponding coarse ladder resistor and keep the current flowing through the fine ladder is relatively an constant. 
     In a further aspect of PGA  500 , a bank of PMOS devices  570  is parallel coupled with the NMOS banks  571 ,  572 ,  573 ,  574 . The PMOS bank  570  has exactly same device count and each device is parallel tied to the corresponding each NMOS device in NMOS banks During the operation, the gate of each PMOS device in the PMOS bank  570  is permanently connected to logic “1”, so that the PMOS devices are turned off all the time during the operation. It seems the PMOS bank  570  is turned off and it is out of circuit operation, however, the back gate leakage currents of the PMOS devices (the leakage current flows through PMOS source/drain parasitic diode from 1.8V back gate NWELL to its source/drain) are used to compensate for the local leakage current generated by the NMOS devices (the leakage current flows from NMOS drain/source to back gate substrate ground). 
     As a result, shown in  FIG. 6 , the back gate leakage current of NMOS parasitic diodes is provided by a local PMOS back-gate leakage current. Note that in  FIG. 6 , only PMOS devices  532 , 533  and NMOS devices  581 ,  582 ,  583 ,  584 , all associated with node  599 , have been drawn for simplicity. The channels of fine ladder MOS resistors experience same amount net current from the beginning to the end along the fine ladder, without being distracted by local leakage current. Therefore the current flowing through the fine ladder is consistently in one direction and it is far away the current level conquered by the leakage current, thus the non-linear scallop described earlier has been eliminated. 
     During the PGA  500  circuit operation, because of OPA virtual ground principle, the voltage potential of the every node of the finer ladder, as well as the two ends of the coarse ladder resistor which is chosen to be shunted with, are all close to VREF=0.9V. Thus, the voltage across the NMOS and PMOS back gate parasitic diodes constantly remains at about 0.9V, the leakage current going through the back gate diodes of either NMOS or PMOS are constant during the operation. The leakage current from PMOS device tied to one node compensates for all nMOS devices tied with the associated node. Note that the PMOS/NMOS back gate leakage is independent of the PMOS/NMOS on or off state, therefore the back gate of the PMOS and NMOS banks in the fine ladder presents a constant leakage current all the time, no matter how many number of the NMOS banks are turned on or turned off. That is, the leakage compensation is independent of the adaptive control operation of NMOS banks. 
       FIG. 7  illustrates a comparison between the prior art gain plot (dashed trace) and the new art gain plot (solid trace) while digital programming code excises from 0000H to 1FFFH. The lower left portion shows the zoom-in plot of the most right portion of  FIG. 6 , the non-linear scallop can be clearly observed in prior art gain plot (red trace) and completely eliminated in new art gain plot (solid trace). 
     Those skilled in the art to which this application relates will appreciate that other and further additions, deletions, substitutions and modifications may be made to the described embodiments.