Abstract:
A frequency demodulation apparatus and a method therefor using an arcsin approximation includes: an analog-to-digital converter (ADC) for sampling a frequency-modulated analog signal to convert the frequency-modulated analog signal into a digital signal, and for outputting the sampled values (S 0 ˜S n ) of the digital signal; a first multiplier for multiplying sinusoidal first and second oscillation signals having a phase difference of 90°, by the sampled value S i (O≦i≦n) respectively, and outputting the products as the signals I i  and Q i ; a low pass filter for low-pass-filtering the signals I i  and Q i  and outputting the low-pass-filtered signals I i ′ and Q i ′; a first de-emphasizer for de-emphasizing the high-frequency components of the signals I i ′ and Q i ′ and outputting the de-emphasized signals I i ″ and Q i ″; a frequency differentiator for delaying the signals I i ″ and Q i ″, multiplying and then subtracting the delayed signals, and outputting the subtraction result as a frequency-demodulated digital signal Z; and a gain corrector for correcting the gain of the signal Z. Efficient frequency demodulation is achieved in a manner especially attractive for SECAM chroma demodulation applications.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to frequency demodulation for demodulating a frequency-modulated signal, and more particularly, to a frequency demodulation apparatus and method for demodulating the signal using an arcsin approximation. 
     2. Description of the Related Art 
     A conventional apparatus for demodulating a frequency-modulated analog signal is disclosed in U.S. Pat. Nos. 5,194,938 and 4,232,268, incorporated herein by reference. The disclosed conventional frequency demodulation apparatus is large and complicated due to the need for an accurate phase locked loop (PLL). In addition, the first conventional frequency demodulation apparatus performs analog frequency demodulation which is greatly affected by process parameters, resulting in increased circuit size. Furthermore, in the case where the conventional frequency demodulation apparatus is employed to demodulate a SECAM (SEquential Couleur Avec Memoire or SEquential Color with Memory) color difference signal, frequencies of sub-carriers for the lines corresponding to first and second color difference signals D b  and D r  are different. For this reason, a PLL for each line must be provided. 
     A second conventional frequency demodulation apparatus has been disclosed in an article entitled “An arctangent type wideband PM/FM demodulator with improved performance”, IEEE TRANSACTION ON CONSUMER ELECTRONICS, Vol. 38, No. 1, pp 5-9, in February 1992, incorporated herein by reference. However, the arctangent approximation used in the second conventional frequency demodulation apparatus results in a large amount of error. Furthermore, the second conventional frequency demodulation apparatus requires a look-up table for the arctangent function in order to attain desired performance, the hardware complexity and manufacturing costs increase. 
     A third conventional frequency demodulation apparatus has been disclosed in an article entitled “A novel FM demodulation scheme”, IEEE TRANSACTION ON CONSUMER ELECTRONICS, Vol. 41, No. 4, pp 1103-1107 in November 1995. This technique requires an additional look up table for calculating a square root function, and, as resolution increases, so too must the size of the look-up table, thus once again, increasing hardware complexity and cost. 
     SUMMARY OF THE INVENTION 
     To address the above limitations, it is a first object of the present invention to provide a frequency demodulation apparatus for performing frequency demodulation using an arcsin approximation. 
     It is a second object of the present invention to provide a frequency demodulation method performed in the frequency demodulation apparatus. 
     It is a third object of the present invention to provide a frequency demodulation apparatus for SECAM (SEquential Couleur Avec Memoire or SEquential Color with Memory) chroma demodulation, which demodulates the frequency of color difference signals transmitted according to the SECAM protocol, using an arcsin approximation. 
     It is a fourth object of the present invention to provide a frequency demodulation method for SECAM chroma demodulation, performed by the frequency demodulation apparatus for the SECAM chroma demodulation. 
     Accordingly, to achieve the above first object, there is provided a frequency demodulation apparatus comprising: an analog-to-digital converter (ADC) for sampling a frequency-modulated analog signal to convert the frequency-modulated analog signal into a digital signal, and outputting the sampled values (S 0 ˜S n ) of the digital signal; a first multiplier for multiplying a sinusoidal first and a second oscillation signals having a phase difference of 90°, by the sampled value S i  (0≦i≦n) respectively, and outputting the products as the signals I i  and Q i ; a low pass filter for low-pass-filtering the signals I i  and Q i  and outputting the low-pass-filtered signals I i ′ and Q i ′; a first de-emphasizer for de-emphasizing the high-frequency components of the signals I i ′ and Q i ′ and outputting the de-emphasized signals I i ″ and Q i ″; a frequency differentiator for delaying the signals I i ″ and Q i ″, multiplying and then subtracting the delayed signals, and outputting the subtraction result as a frequency-demodulated digital signal Z; a gain corrector for correcting the gain of the signal Z; and an oscillator for outputting the first and second oscillation signals. 
     To achieve the second object, there is provided a frequency demodulation method comprising the steps of: (a) sampling a frequency-modulated analog signal to calculate sampled values S 0 ˜S n ; (b) multiplying first and second oscillation signals that are sinusoidal with a phase difference of 90° by the sampled value S i  (0≦i≦n) respectively and obtaining the products I i  and Q i ; (c) low-pass-filtering the signals I i  and Q i  to obtain the low-pass-filtered signals I i ′ and Q i ′; (d) de-emphasizing the high-frequency components of the signals I i ′ and Q i ′ to obtain the de-emphasized signals I i ″ and Q i ″; (e) delaying the signals I i ″ and Q i ″, multiplying and then subtracting the delayed signals to obtain a frequency-demodulated digital signal Z; and (f) correcting the gain of the frequency-demodulated digital signal Z. 
     To achieve the third object, i.e., for the SECAM chroma demodulation, the frequency demodulation apparatus further comprises: a first selector for selectively outputting the signals I i ″ and Q i ″ or the signals I i ′ and Q i ′ in response to a selection signal; a gain adjustor for determining whether the amplitude of the signal I i ′ maintains a predetermined value, adjusting the gain of the signal I i ′ in response to the determination result, and outputting the gain-adjusted signal to the first multiplier; a line checking unit for checking whether the current line is an even line or odd line using the signal Z, the selection signal and the delayed signals, and outputting the checking result; a second de-emphasizer for de-emphasizing the low-frequency component of the signal output from the gain corrector, and outputting the de-emphasized result; and a color difference signal reproducer for differentiating and outputting the signals for the previous line among the signals output from the second de-emphasizer as a frequency-demodulated first color difference signal and for differentiating and outputting the signals for the current line among the signals output from the second de-emphasizer as a frequency-demodulated second color difference signal, in response to the line checking result of the line checking unit, wherein the analog signal includes the frequency-modulated first or second color difference signal, the color difference signal transmitted being loaded on the odd or even line, the frequency differentiator calculates the signal Z using the result selected by the first selector, (instead of using the signals I i ″ and Q i ″), the first multiplier multiplies the signals I i  and Q i  by the gain adjusted by the gain adjustor and outputs the products to the low pass filter, the gain corrector corrects the gain and offset of the signal Z in response to the check result of the line checking unit and outputs the corrected result to the second de-emphasizer, and the selection signal is generated during the interval in which a color burst signal exists in the analog signal. 
     To achieve the fourth object, there is provided a frequency demodulation method for the SECAM chroma demodulation, comprising the steps of: (a) sampling an analog composite video baseband signal with a frequency-modulated first or second color difference signal to obtain sampled values S 0 ˜S n ; (b) multiplying third and fourth oscillation signals that are sinusoidal with a phase difference of 90° by the sampled value S i  (0≦i≦n) respectively and obtaining the products I i  and Q i ; (c) low-pass-filtering the signals I i  and Q i  to obtain the low-pass-filtered signals I i ′ and Q i ′; (d) delaying the signals I i ′ and Q i ′, multiplying and then subtracting the delayed signals to obtain a signal Z; (e) determining whether the current line is an even line or odd line using the signal Z, the selection signal and the delayed signals; (f) determining the signals Z for the previous line as a frequency-demodulated first color difference signal if the current line is an even line; and (g) determining the signals Z for the current line as a frequency-demodulated second color difference signal if the current line is an odd line, wherein the frequency-modulated first and second color difference signals are loaded on the odd or even line and transmitted. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The foregoing and other objects, features and advantages of the invention will be apparent from the more particular description of preferred embodiments of the invention, as illustrated in the accompanying drawings in which like reference characters refer to the same parts throughout the different views. The drawings are not necessarily to scale, emphasis instead being placed upon illustrating the principles of the invention. 
     FIG. 1 is a block diagram of a frequency demodulation apparatus according to a preferred embodiment of the present invention. 
     FIG. 2 is a flowchart illustrating a frequency demodulation method according to the present invention, performed in the apparatus shown in FIG.  1 . 
     FIG. 3 is a diagram illustrating the principle of differentiation using an arcsin approximation. 
     FIG. 4 is a circuit diagram of an embodiment of the first frequency differentiator shown in FIG. 1 according to the present invention. 
     FIG. 5 is a block diagram of a frequency demodulation apparatus according to another embodiment of the present invention. 
     FIG. 6 is a flowchart illustrating a frequency demodulation method performed in the apparatus shown in FIG. 5 according to the present invention. 
     FIG. 7 is a circuit diagram of a frequency demodulation apparatus for SECAM (SEquential Couleur Avec Memoire or SEquential Color with Memory) chroma demodulation according to a first embodiment of the present invention. 
     FIG. 8 is a flowchart illustrating a frequency demodulation method for the SECAM chroma demodulation performed in the apparatus of FIG. 7 according to the present invention. 
     FIG. 9 is a circuit diagram of a frequency demodulation apparatus for SECAM chroma demodulation according to a second embodiment of the present invention. 
     FIG. 10 is a flowchart illustrating a frequency demodulation method for the SECAM chroma demodulation performed in the apparatus of FIG. 9 according to the present invention. 
     FIG. 11 is a circuit diagram of a frequency demodulation apparatus for SECAM chroma demodulation according to a third embodiment of the present invention. 
     FIG. 12 is a circuit diagram of a frequency demodulation apparatus for SECAM chroma demodulation according to a fourth embodiment of the present invention. 
     FIG. 13 is a block diagram of an example of the first, second, third or fourth line checking unit of FIG. 7,  9 ,  11  or  12  according to an embodiment of the present invention. 
     FIG. 14 is a circuit diagram of an example of the frequency detector of FIG. 13 according to an embodiment of the present invention. 
     FIG. 15 is a circuit diagram of an embodiment of the first or second gain and offset controller of FIG. 7 or  9 , respectively, according to the embodiment of the present invention. 
     FIG. 16 is a circuit diagram of an example of the third or fourth gain corrector of FIG. 11 or  12  according to the embodiment of the present invention. 
     FIG. 17 is a circuit diagram of an example of the first, second, third or fourth color difference signal reproducer according to the embodiment of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Hereinafter, the structure and operation of a frequency demodulation apparatus according to the present invention, and a frequency demodulation method thereof will be described with reference to the appended drawings. 
     Referring to FIG. 1, a frequency demodulation apparatus according to a preferred embodiment of the present invention includes a first analog-to-digital converter (ADC)  10 , a first oscillator  12 , a multiplier  14 , a first low pass filter (LPF)  16 , a first de-emphasizer  18 , a first frequency differentiator  20 , and a first gain corrector  22 . 
     FIG. 2 is a flowchart illustrating a frequency demodulation method according to the present invention performed in the apparatus shown in FIG. 1, in which a digital signal converted from a frequency-modulated analog signal is frequency-demodulated using an arcsin approximation (steps  30  through  38 ). 
     The first ADC  10  of FIG. 1 receives through an input port IN 1  an analog signal [X(t)] containing a frequency-modulated message signal [M(t)], which is expressed below as Equation (1), samples the input analog signal [X(t)] in response to a sampling clock signal CK s  to convert it into a digital signal, and outputs sampled values S 0 ˜S n , i.e., the digital signal [X(nT)], expressed below as Equation 2, to the multiplier  14  (step  30 ).                X        (   t   )       =       A        (   t   )       *     cos        (       2      π                   f   0        t     +       k   f            ∫   0   t            M     (   T   )               T             )                 (   1   )                                
     In Equation (1), f o  indicates the carrier frequency of the analog signal [X(t)], and k f  indicates the frequency deviation.                X        (   nT   )       =     Y   +       A        (   nT   )       *     cos        (       2      π                   f   0        nT     +       k   f            ∑     p   =   0     n                     M        (   p   )             )                   (   2   )                                
     Following step  30 , the multiplier  14  multiplies first and second oscillation signals G i1  and G i2  output from the first oscillator  12 , expressed below as Equation (3), by the sampled values S i  (0≦i≦n), respectively, and outputs the products I i  and Q i  expressed below as Equation (4) to the first LPF (step  32 ). 
     
       
           G   i1 =sin(2 πf   SC   iT ) G   i2 =cos(2 πf   SC iT)  (3) 
       
     
     In Equation (3), G i1  and G i2  represent sinusoidal waves with a 90° phase difference and have a predetermined free running frequency f SC .                      I   i     =                  S   i     *     G   i1                   =                  [     Y   +       A        (   iT   )       *     cos        (       2      π                   f   0        iT     +       k   f            ∑     p   =   0     i          M        (   p   )             )           ]     *     sin        (     2      π                   f   sc        iT     )                       Q   i     =                  S   i     *     G   i2                   =                  [     Y   +       A        (   iT   )       *     cos        (       2      π                   f   0        iT     +       k   f            ∑     p   =   0     i          M        (   p   )             )           ]     *     cos        (     2      π                   f   sc        iT     )                       (   4   )                                
     In the Equation (4), I represents the “in phase” component and Q represents the “quadrature” component. 
     Following step  32 , the first LPF  16  low-pass-filters the signals I i  and Q i  output from the multiplier  14  and outputs the filtered results I i ′ and Q i ′, expressed as Equation (5), to the first de-emphasizer  18  (step  34 ).                  I   i   ′     =         A        (   iT   )       2     *     sin        [       2        π        (       f   sc     -     f   0       )          iT     -       k   f            ∑     p   =   0     i          M        (   p   )             ]                
            Q   i   ′     =         A        (   iT   )       2     *     cos        [       2        π        (       f   0     -     f   sc       )          iT     +       k   f            ∑     p   =   0     i          M        (   p   )             ]                   (   5   )                                
     Here, if the free running frequency f SC  of the respective first and second oscillation signals is f o , that is, if the free running frequency f SC  is identified with the carrier frequency f o  of the frequency-modulated analog signal [M(t)], then Equation (5) can be expressed as Equation (6).                  I   i   ′     =       -       A        (   iT   )       2       *     sin        [       k   f            ∑     p   =   0     i          M        (   p   )           ]                
            Q   i   ′     =         A        (   iT   )       2     *     cos        [       k   f            ∑     p   =   0     i          M        (   p   )           ]                   (   6   )                                
     Following step  34 , the first de-emphasizer  18  de-emphasizes high frequency components of the signals I i ′ and Q i ′ output from the first LPF  16 , and outputs the de-emphasized results I i ″ and Q i ″, expressed as Equation (7), to the first frequency differentiator  20  (step  36 ).                  I   i   ″     =     B   *     sin        [       k   f            ∑     p   =   0     i          M        (   p   )           ]                
            Q   i   ″     =     B   *     cos        [       k   f            ∑     p   =   0     i          M        (   p   )           ]                   (   7   )                                
     In Equation (7), B is a value obtained by normalizing            A        (   iT   )       2     .                          
     Following step  36 , the first frequency differentiator  20  differentiates signals I i ″ and Q i ″ output from the first de-emphasizer  18  using an arcsin approximation and outputs a frequency-modulated digital signal Z, which is the result of the differentiation, to the first gain corrector  22  (step  38 ). 
     Hereinafter, the differentiation using the arcsin approximation will be explained. Here, for utilizing the arcsin approximation, it is assumed that the width        [       k   f            ∑     p   =   0     i          M        (   p   )           ]                          
     in the frequency deviation of the integrated message signal based on the frequency of the sampling clock signal CK S  is not great. 
     FIG. 3 is a diagram illustrating the principle of the differentiation using the arcsin approximation, in which the X-axis indicates the in-phase signal I i ″ and the Y-axis indicates the quadrature signal Q″. 
     The sampled values I i ″ and Q i ″ expressed by Equation (7) and the neighboring sampled values I i+1 ′ and Q i+1 ″ are located on the same circumference, i.e. contour, of a circle as shown in FIG.  3 . Thus, assuming that          α   =         k   f            ∑     p   =   0     i            M        (   p   )                     and                 β         =       k   f            ∑     p   =   0       i   +   1            M        (   p   )               ,                          
     the message signal can be extracted by Equation (8) by calculating the angle difference (β−α) between two neighboring sampled values.                          I   i   ″     *     Q     i   +   1     ″       -       I     i   +   1     ″     *     Q   i   ″         =                  B   2     *     {       sin                   β   ·   cos                   α     -     sin                   α   ·   cos                   β       }                   =                    B   2     *     sin        (     β   -   α     )         =       B   2          sin        [       k   f          M        (     i   +   1     )         ]                         (   8   )                                
     In Equation (8), if k f  M(i+1) is very small, sin[k f  M(i+1)] approximates k f  M(i+1), which is referred to herein as the arcsin approximation. Here, a frequency-modulated digital signal [B 2 k f M(i)](Z) is expressed as Equation (9). 
     
       
           B   2   k   f   M ( i +1)= I   i   ″*Q   i+1   ″−I   i−1   ″*Q   i ″  (9) 
       
     
     Thus, the first frequency differentiator  20  of FIG. 1 performs the function of Equation (9) obtained through the arcsin approximation in order to differentiate the signals I i ″ and Q i ″. That is, the first frequency differentiator  20  performs the multiplications and subtraction on signals, I i ″ and Q i ″, I i+1 ″ and Q i+1 ″ in order to obtain the frequency-demodulated digital signal [B 2 k f M(i)]. The structure and operation of the first frequency differentiator  20  that performs the function of Equation (9) will be explained. 
     FIG. 4 is a circuit diagram of an embodiment of the first frequency differentiator  20  shown in FIG.  1 . The first frequency differentiator  20  includes first, second, third and fourth delays  42 ,  44 ,  46  and  48 , multipliers  50  and  52  and a subtractor  54 . 
     In FIG. 4, the first delay  42  delays the signal I i ″ input through an input port IN 2  from the first de-emphasizer  18  and outputs the delayed result U 2  to the second delay  44 . The second delay  44  delays the output of the first delay  42  and outputs the delayed result U 1  to the multiplier  50 . 
     Similarly, the third delay  46  delays the signal Q i ″ input through an input port IN 3  from the first de-emphasizer  18  and outputs the delayed result V 2  to the fourth delay  48 . The fourth delay  48  delays the output of the third delay  46  and outputs the delayed result V 1  to the multiplier  52 . 
     Here, the multiplier  50  multiplies the output U 1  of the second delay  44  by the output V 2  of the third delay  46 , and outputs the product to the subtractor  54 . The multiplier  52  multiplies the output U 2  of the first delay  42  by the output V 1  of the fourth delay  48  and outputs the product to the subtractor  54 . The subtractor  54  subtracts the product U 1 *V 2  of the multiplier  50  from the product V 1 *U 2  of the multiplier  52 , and outputs the subtraction result as the frequency-demodulated digital signal [M(i+1)], i.e., Z 
     Each of the first, second, third and fourth delays  42 ,  44 ,  46  and  48  may be implemented, for example, by a D flip-flop which latches data in response to the sampling clock signal CK S . 
     Next, the first gain corrector  22  of FIG. 1 adjusts the gain of the frequency-demodulated digital signal [B 2 k f M(i)] and outputs the gain-adjusted message signal [M(i)] via an output port OUT. 
     FIG. 5 is a block diagram of a frequency demodulation apparatus according to another embodiment of the present invention. The frequency demodulation apparatus includes a second ADC  56 , a second de-emphasizer  58 , a second oscillator  60 , a multiplier  62 , a second LPF  64 , a second frequency differentiator  66  and a second gain corrector  68 . 
     FIG. 6 is a flowchart illustrating a frequency demodulation method executed by the apparatus of FIG. 5, in which the digital signal converted from the frequency-modulated analog signal is frequency-demodulated using an arcsin approximation (steps  70  through  78 ). 
     The frequency modulation apparatus of FIG. 5 has the same structure and function as that of FIG. 1 except for the arrangement of the second de-emphasizer  58 , and performs the same operation as that of FIG.  1 . That is, while the first de-emphasizer  18  of FIG. 1 is arranged between the first LPF  16  and the first frequency differentiator  20 , the second de-emphasizer  58  of FIG. 5 is placed between the second ADC  56  and the multiplier  62 . In FIG. 5, the second ADC  56 , the second oscillator  60 , the second LPF  64  and the second gain corrector  68  correspond to the first ADC  10 , the first oscillator  12 , the first LPF  16  and the first gain corrector  22  of FIG. 1, respectively. Thus, step  70  of FIG. 6 corresponds to step  30  of FIG.  2 . 
     Following step  70 , the second de-emphasizer  58  of FIG. 5 de-emphasizes high-frequency components of the sampled valued S i  output from the second ADC  56 , and outputs the de-emphasized result to the multiplier  62  (step  72 ). Then, the multiplier  62  multiplies by first and second oscillation signals the de-emphasized result output from the second de-emphasizer  58 , instead of the sampled value S i , and outputs the products, signals I j  and Q j  (0≦j≦n) to the second LPF  64  (step  74 ). Here, if in Equation (4), the subscript i is replaced by the subscript j and the amplitude [A(j)] is replaced by the normalized value B expressed in the Equation (7), the signals I j  and Q j  are actually the same as I i  and Q i  of Equation (4). 
     Following step  74 , the products I j  and Q j  output from the multiplier  62  are low-pass-filtered in the second LPF  64  (step  76 ). Then, the second frequency differentiator  66  differentiates the signals I j′ and Q   j ′, instead of the signals I i ″ and Q i ″ as in the FIG. 1 embodiment described above, using the above-explained arcsin approximation, and outputs the result of differentiation, that is, the frequency-demodulated digital signal [B 2 k f M(j)], to the second gain corrector  68  (step  78 ). That is, the second frequency differentiator  66  performs the same operation as in the first frequency differentiator  20  of FIG. 1, except that the signals input to the first and second frequency differentiators  20  and  66  are different. 
     Hereinafter, the structure and operation of a frequency modulation apparatus for SECAM chroma demodulation, corresponding to the frequency modulation apparatus of FIGS. 1 or  5 , which demodulates color difference signals transmitted in the SECAM fashion, and a frequency demodulation method therefor will be described. 
     SECAM is a video broadcast standard developed in France, in which two color difference signals D b  and D r  are alternately transmitted through even and odd lines, respectively, so as to prevent crosstalk between the chroma signals. That is, unlike the NTSC (National Television System Committee) and PAL (Phase Alternation by Line), the red-yellow R-Y component (D r ) loaded onto either an odd or an even line, and the blue-yellow B-Y component (D b ) loaded onto the following line having D r  are transmitted to the receiving part. Here, a luminance component is included in every line in the transmission. In the SECAM fashion, frequency modulation is used for the transmission of the color difference signals, and the two color difference signals each have an independent sub-carrier wave. 
     Assuming that the frequency-modulated analog signal [X(t)] of the Equation (1) is an analog CVBS (Composite Video Baseband Signal) with a color bust signal, the message signal [M(i)] is a first or second color difference signal D b (i) or D r (i), and f o  is the sub-carrier frequency f ob  or f or  of an analog CVBS having the first or second color difference signal D b (i) or D r (i), the above description with reference to FIG. 1 or  5  can be applied to a frequency demodulation apparatus for SECAM chroma demodulation. Here, in a color burst interval where the color burst signal exists, the amplitude [A(t)] of the CVBS becomes a constant and the first or second color difference signal D b (i) or D r (i) does not exist. Also, in a video active interval where the color burst signal does not exist, the amplitude [A(t)] becomes a function of time. Here, f ob  and f or  may be 4.25 MHz(±2 kHz) and 4.4 MHz (±2 kHz), respectively. 
     FIG. 7 is a circuit diagram of a frequency demodulation apparatus for SECAM chroma demodulation according to a first embodiment of the present invention. The frequency demodulation apparatus includes a third ADC  90 , a third oscillator  92 , a multiplier  94 , a third LPF  96 , a first gain adjustor  98 , a third de-emphasizer  100 , a multiplexer  102 , a third frequency differentiator  104 , a first line checking unit  106 , a first color difference signal reproducer  108 , a first gain and offset controller  110 , a fourth de-emphasizer  112  and a first coring unit  114 . 
     FIG. 8 is a flowchart illustrating a frequency demodulation method for a SECAM chroma demodulation according to the present invention, performed in the apparatus of FIG. 7, in which a digital CVBS converted from an analog CVBS is demodulated into a signal Z (steps  120  through  132 ) and the signal Z is divided in lines to determine first and second color difference signals (steps  134  through  138 ). 
     The third ADC  90  of FIG. 7 receives an analog signal CVBS with a frequency-modulated first or second color difference signal D b (i) or D r (i), samples the input analog CVBS in response to a sampling clock signal CK S , in order to convert the analog signal into a digital signal, and outputs digital sampled values S 0 ˜S n  to the multiplier  94  (step  120 ). 
     Then, the multiplier  94  multiplies the first and second oscillation signals output from the third oscillator  92  by the sampled values S i  (0≦i≦n) output from the third ADC  90 , respectively, and outputs the products I i  and Q i  to the third LPF  96  (step  122 ). To this end, the third oscillator  92  outputs the first and second oscillation signals expressed by Equation (3) having the free running frequency f SC  as in the same manner as in the first oscillator  12  of FIG.  1 . Here, the frequency of the above-mentioned sampling clock signal CK S  may be 13.5 MHz and the free running frequency f SC  may be 4.33 MHz, wherein the frequency of the sampling clock signal CK S  and the free running frequency f SC  must satisfy the Nyquist rate. 
     Following step  122 , the third LPF  96  low-pass-filters the signals I i  and Q i  and outputs the low-pass-filtered signals I i ′ and Q i ′ to the third de-emphasizer  100  (step  124 ). 
     Here, the first gain adjustor  98  may be further provided between the third LPF  96  and the multiplier  94 , checks whether the amplitude of the signal I i ′ output from the third LPF  96  maintains a predetermined value during the color burst interval, and outputs to the multiplier  94  a gain determined in response to the check result. Thus, the multiplier  94  multiplies the sampled values by the gain determined in the first gain adjustor  98 , and then by the respective first and second oscillation signals, to output signals I i  and Q i . That is, because there is a high probability that the level of the first or second color difference signals is low if the level of the color burst signal is less than a predetermined level, it is necessary for the first gain adjustor  98  to keep the level of the low-pass-filtered result constant. 
     Following step  124 , the third de-emphasizer  100  de-emphasizes high-frequency components of the signals I i ′ and Q i ′ and outputs the de-emphasized signals I i ″ and Q i ″ to the multiplexer  102  (step  126 ). When the first and the second color different signals are transmitted using the SECAM fashion to the receiving part from a transmitting part, high-frequency components of the color difference signals are very susceptible to noise, so the signal-to-noise distortion ratio (SNDR) may be lowered. Thus, at the transmitting part pre-emphasis is performed to increase the amplitude of the high-frequency component. Also, at the receiving part the high-frequency components of the signals I i ′ and Q i ′ are de-emphasized using the third de-emphasizer  100  as mentioned above when the color difference signals are frequency-demodulated. The third de-emphasizer may comprise, for example, a cloche filter or a bell filter. 
     Following step  126 , it is determined whether the current interval is the color burst interval (step  128 ). If so, the third frequency differentiator  104  obtains a signal Z using the signals I i ′ and Q i ′ output from the third LPF  96  (step  130 ). Otherwise, the third frequency differentiator  104  obtains the signal Z using the signals I i ″ and Q i ″ output from the third de-emphasizer  100  (step  132 ). To this end, a selection signal S is generated such that the multiplexer  102  selects the outputs I i ′ and Q i ′ of the third LPF  96  during the color burst interval and the outputs I i ″ and Q i ″ of the third de-emphasizer  100  during the video active interval. For example, the selection signal S can be generated in a controller (not shown) depending on whether a color burst signal is present, for example, such that it maintains a high level during the color burst interval and a low level during the video active interval. 
     The third frequency differentiator  104  performs the same operation as in the first frequency differentiator  20  of FIG. 1 or the second frequency differentiator  66  of FIG.  5 . That is, the Equation (8) can be expressed as Equation (10).                          X   i     *     Y     i   +   1         -       X     i   +   1       *     Y   i         =                  B   2     *     {       sin                   β   ·   cos                   α     -     sin                   α   ·   cos                   β       }                   =                  B   2     *     sin        (     β   -   α     )                       =                  B   2     *     sin        [       k   f            D   r          (     i   +   1     )         ]                     or                                             B   2     *     sin        [       k   f            D   b          (     i   +   1     )         ]                                  (   10   )                                
     In Equation (10), X i  and Y i  correspond to the signals I i ″ and Q i ″ of Equation (8), α corresponds to            k   f            ∑     p   =   0     i              D   r          (   p   )                     or                   k   f            ∑     p   =   0     i            D   b          (   p   )               ,                          
     and β corresponds to          k   f            ∑     p   =   0       i   +   1                D   r          (   p   )                     or                   k   f            ∑     p   =   0       i   +   1                D   b          (   p   )       .                                  
     As described above, sin(β−α) expressed in the Equation (10) approximates to β−α using the aresin approximation. Actually, the maximum width of the frequency deviation of the first or second color difference signal information integrated based on the frequency of the sampling clock signal CK S  used in the third ADC  90  is less than 500 kHz, so that β−α becomes a maximum of 13.3° (13.5 MHz:500kHz=360°:13.3°). Because the interval from 0˜13.3° in the sine function is nearly linear, it is understood that the arcsin approximation according to the present invention can be applied to the actual circumstances. For example, if the third de-emphasizer  100  normalizes the amplitude [A(nT)] to B, B 2 k f D r (i+1) or B 2 k f D b (i+1) approximate X i *Y i+1 −X i+1 *Y i . Thus, the third frequency differentiator  104  may be implemented by the circuit of FIG. 4, and the signal Z corresponds to B 2k   f D r (i+1) or B 2 k f D b (i+1). 
     After the step  130  or  132 , the first line checking unit  106  decides whether the current  1   5  line is an even line or an odd line (step  134 ). If the current line is an even line, the signal Z on the previous line is decided as a frequency-demodulated first color difference signal [D b ′] (step  136 ). 
     However, if the current line is an odd line, the signal Z on the current line is decided as a frequency-demodulated second color difference signal [D r ′] (step  138 ). To this end, the first line checking unit  106  checks whether the current line is an odd line or an even line in response to the signal Z, the selection signal S and delayed results U 1 /U 2  and V 1 /V 2 , and outputs the check result LD to the first color difference signal reproducer  108 . The first color difference signal reproducer  108  extracts the signal Z on the previous line and the signal Z on the current line as the frequency-demodulated first and second color difference signals [D b ′] and [D r ′], respectively, in response to the logic level of the check result LD, and outputs the first and second color difference signals [D b ] and [D r ′] to the first coring unit  114 . 
     When Equation (5) is approximated to Equation (6), f SC −f ob  or f SC −f or  is neglected. However, f SC −f ob  or f SC −f or  should not be neglected in the actual situation. If f SC −f ob  or f SC −f or  is neglected, the demodulated color difference signals may have an offset. Thus, the first gain and offset controller  110  may be used to compensate for the offset caused by the above-mentioned approximation, so as to adjust and to compensate for the gain and offset of the signal Z in response to the check result LD, and outputs the gain-adjusted and offset-compensated result to the fourth de-emphasizer  112 . In addition, the first gain and offset controller  110  operates as the first gain corrector  22  of FIG. 1 or the second gain corrector  68  of FIG.  5 . 
     In the SECAM fashion, at the transmitting part from which color difference signals are transmitted, pre-emphasis is performed on respective low-frequency components of the first and second color different signals in order to improve noise immunity. Thus, the frequency demodulation apparatus of the receiving part includes the fourth de-emphasizer  112  as shown in FIG.  7 . The fourth de-emphasizer  112  de-emphasizes the low-frequency component of the gain- and offset-controlled signal output from the first gain and offset controller  110 , and outputs the de-emphasized signal Z′″ to the first color difference signal reproducer  108 . 
     In a preferred embodiment, the first coring unit  114  is provided. The first coring unit  104  detects as a noise a small change in the amplitudes of the frequency-demodulated first and second color difference signals [D b ′] and [D r ′] separated in the first color difference signal reproducer  108 , changes the level of the difference signals to a predetermined level, and outputs final first and second color difference signals D b  and D r  having the changed levels. 
     FIG. 9 is a circuit diagram of a frequency demodulation apparatus for the SECAM chroma demodulation according to a second embodiment of the present invention. The frequency demodulation apparatus of FIG. 9 comprises a fourth ADC  150 , a fourth oscillator  152 , a multiplier  154 , a fourth LPF  156 , a second gain adjustor  158 , a fifth de-emphasizer  160 , a multiplexer  162 , a fourth frequency differentiator  164 , a second line checking unit  166 , a second color difference signal reproducer  168 , a second gain and offset controller  170 , a sixth de-emphasizer  172  and a second coring unit  174 . 
     FIG. 10 is a flowchart illustrating a frequency modulation method for SECAM chroma demodulation according to the present invention, performed in the apparatus of FIG. 9, in which a digital CVBS converted from an analog CVBS is demodulated to extract a signal Z (steps  180  through  192 ) and the obtained signal Z is divided across lines to determine color difference signals (steps  194  through  198 ). 
     The frequency modulation apparatus of FIG. 9 has the same structure as that of FIG. 7 except that the fifth de-emphasizer  160  and the multiplexer  162  are arranged between the fourth ADC  150  and the multiplier  154 , and performs the same operation as that of FIG.  7 . That is, the fourth ADC  150 , the fourth oscillator  152 , the second gain adjustor  158 , the fourth frequency differentiator  164 , the second line checking unit  166 , the second color difference signal reproducer  168 , the second gain and offset controller  170 , the sixth de-emphasizer  172  and the second coring unit  174  of FIG. 9 correspond to the third ADC  90 , the third oscillator  92 , the first gain adjustor  98 , the third frequency differentiator  104 , the first line checking unit  106 , the first color difference signal reproducer  108 , the first gain and offset controller  110 , the fourth de-emphasizer  112  and the first coring unit  114  of FIG. 7, respectively, and perform the same operations. Thus, step  180 , and steps  194  through  198  correspond to step  120 , and steps  134  through  138 , respectively. 
     Following step  180 , the fifth de-emphasizer  160  de-emphasizes the high-frequency component of the sampled value output from the fourth ADC  150 , and outputs the de-emphasized result to the multiplexer  162  (step  182 ). Next it is determined if the current interval is a color burst interval (step  184 ). If so, the multiplier  154  multiplies the sampled value by the first and second oscillation signals respectively (step  186 ). Otherwise, the multiplier  154  multiplies the de-emphasized result output from the fifth de-emphasizer  160  by the first and second oscillation signals respectively, in a video active interval (step  188 ). To this end, a selection signal S is generated such that the multiplexer  162  selects the sampled value output from the fourth ADC  150  in the color burst interval and the output of the fifth de-emphasizer  160  in the video active interval. 
     Following step  186  or  188 , the fourth LPF  156  low-pass-filters the products I i  and Q i  of the multiplier  154  and outputs the filtered results I i ′ and Q i ′ to the fourth frequency differentiator  164  (step  190 ). Following step  190 , the fourth frequency differentiator  164  which performs the same operation as by the third frequency differentiator  104  of FIG. 7 differentiates the filtered result output from the fourth LPF  156  using the above-described arcsin approximation and outputs the differentiated result as a signal Z (step  192 ). 
     FIG. 11 is a circuit diagram of a frequency demodulation apparatus for SECAM chroma demodulation according to a third embodiment of the present invention. The frequency demodulation apparatus of FIG. 11 comprises a fifth ADC  290 , a fifth oscillator  292 , a multiplier  294 , a fifth LPF  296 , a third gain adjustor  298 , a seventh de-emphasizer  300 , a multiplexer  302 , a fifth frequency differentiator  304 , a third line checking unit  306 , a third color difference signal reproducer  308 , a third gain corrector  310 , an eighth de-emphasizer  312 , a third coring unit  314  and a first frequency tracking unit  316 . 
     The structure and operation of the frequency demodulation apparatus of FIG. 11 are the same as those of the frequency demodulation apparatus of FIG. 7, except that the first frequency tracking unit  316  is further provided so as to control the oscillation frequency of the fifth oscillator  292 , the third gain corrector  310  is provided instead of the first gain and offset controller  110  of FIG. 7, and the line checking method of the third line checking unit  306  is different from that of the first line checking unit  106  of FIG.  7 . That is, the fifth ADC  290 , the multiplier  294 , the fifth LPF  296 , the third gain adjustor  298 , the seventh de-emphasizer  300 , the multiplexer  302 , the fifth frequency differentiator  304 , the third color difference signal reproducer  308 , the eighth de-emphasizer  312  and the third coring unit  314  of FIG. 11 correspond to the third ADC  90 , the multiplier  94 , the third LPF  96 , the first gain adjustor  98 , the third de-emphasizer  100 , the multiplexer  102 , the third frequency differentiator  104 , the first color difference signal reproducer  108 , the fourth de-emphasizer  112  and the first coring unit  114  of FIG. 7, respectively, and perform the same operations. Thus, the frequency demodulation method illustrated with reference to FIG. 8 can be applied to the apparatus of FIG.  11 . 
     As described above, in the frequency demodulation apparatus of FIG. 11, the first and second oscillation signals do not oscillate independently, unlike the apparatus of FIG.  7 . That is, the first frequency tracking unit  316  tracks the sub-carrier frequency f ob  or f or  of the analog CVBS in response to the check result LD′ output from the third line checking unit  306 , and outputs the track results to the fifth oscillator  292 . Here, the fifth oscillator  292  does not have its own free running frequency and outputs the first and second oscillation signals to the multiplier  294 , the frequencies of which are determined in response to the result tracked by the first frequency tracking unit  316 . 
     Also, unlike the first gain and offset controller  110  of FIG. 7 which controls both the gain and the offset of the signal Z, the third gain corrector  310  of FIG. 11 adjusts only the gain of the signal Z. That is, the third gain corrector  310  adjusts the gain of the signal Z in response to the check result LD′, and outputs the adjust result to the third color difference signal reproducer  308  via the eighth de-emphasizer  312 . Here, the third gain corrector  310  acts as the first gain corrector  22  of FIG. 1 or the second gain corrector  68  of FIG.  5 . 
     Here, the third line checking unit  306  has the same function as that of the first line checking unit  106  of FIG. 7, but its line checking method is different from that of first line checking unit  106 . That is, the first or third line checking unit  106  or  306  checks whether the current line is an odd line or even line by using the signal Z selection signal S and the delayed results U 1 /U 2  and V 1 /V 2  and outputs the check result LD′. However, the first line checking unit  106  of FIG.  7  and the third line checking unit  306  of FIG. 11 compare the frequency of a color burst signal with predetermined frequencies, which will be described below. For example, the predetermined frequency used to check the line in the first line checking unit  106  is 4.33 MHz, but that for the third line checking unit  306  is 4.25 MHz. 
     FIG. 12 is a block diagram of a frequency demodulation apparatus for SECAM chroma demodulation according to a fourth embodiment of the present invention. The frequency demodulation apparatus of FIG. 12 comprises a sixth ADC  330 , a sixth oscillator  332 , a multiplier  334 , a sixth LPF  336 , a fourth gain adjustor  338 , a ninth de-emphasizer  340 , a multiplexer  342 , a sixth frequency differentiator  344 , a fourth line checking unit  346 , a fourth color difference signal reproducer  348 , a fourth gain corrector  350 , a tenth de-emphasizer  352 , a fourth coring unit  354  and a second frequency tracking unit  356 . 
     The frequency demodulation apparatus of FIG. 12 performs the same operation as that of FIG. 9, except the frequency demodulation apparatus of FIG. 12 further comprises the second frequency tracking unit  356  for controlling the oscillation frequency of the sixth oscillator  332 , and the fourth gain corrector  350  is adopted instead of the second gain and offset controller  170 , and the line checking method of the fourth line checking unit  346  is different from that of the second line checking unit  166 . That is, the sixth ADC  330 , the multiplier  334 , the sixth LPF  336 , the fourth gain adjustor  338 , the ninth de-emphasizer  340 , the multiplexer  342 , the sixth frequency differentiator  344 , the fourth color difference signal reproducer  348 , the tenth de-emphasizer  352  and the fourth coring unit  354  shown in FIG. 12 correspond to the fourth ADC  150 , the multiplier  154 , the fourth LPF  156 , the second gain adjustor  158 , the fifth de-emphasizer  160 , the multiplexer  162 , the fourth frequency differentiator  164 , the second color difference signal reproducer  168 , the sixth de-emphasizer  172  and the second coring unit  174  shown in FIG. 9, respectively, and perform the same operations. 
     Also, unlike the frequency demodulation apparatus of FIG. 11 in which the seventh de-emphasizer  300  and the multiplexer  302  are arranged between the fifth LPF  290  and the fifth frequency differentiator  304 , the ninth de-emphasizer  340  and the multiplexer  342  are arranged between the sixth ADC  330  and the multiplier  334  in the frequency demodulation apparatus of FIG.  12 . Except for these distinctions, the frequency reproduction apparatus of FIG. 12 performs substantially the same operation as the frequency demodulation apparatus of FIG.  11 . That is, the sixth oscillator  332 , the second frequency tracking unit  356 , the fourth gain corrector  350  and the fourth line checking unit  346  of FIG. 12 correspond to the fifth oscillator  292 , the first frequency tracking unit  316 , the third gain corrector  310  and the third line detecting unit  306  of FIG. 11, respectively. Thus, the frequency demodulation. method illustrated in FIG. 10 can be applied to the frequency demodulation apparatus of FIG.  12 . 
     Hereinafter, the structure and operation of the first, second, third or fourth line checking unit  106 ,  166 ,  306  or  346  according to an embodiment of the present invention will be described. 
     FIG. 13 is a block diagram showing an example of the first, second, third or fourth line checking unit  106 ,  166 ,  306  or  346 . The line checking unit of FIG. 13 comprises a flip-flop  400 , a frequency detector  402  and a comparator  404 . 
     In the line checking unit of FIG. 13, the flip-flop  400 , a latch, receives the signal Z through a data input port D and a selection signal S through a clock port CK, and outputs the latched Z signal ZD through a positive output port Q to the frequency detector  402 . Here, the frequency detector  402  detects the frequency of a color burst signal using the delayed result U 1 /U 2  or V 1 /V 2  input from the frequency differentiator of FIG.  4  and the latched Z signal ZD, and outputs a detected frequency f CB  to the comparator  404 . The structure of the frequency detector  402  of FIG.  13  and its operation will now be described in detail with reference to FIG.  14 . 
     FIG. 14 is a circuit diagram showing an example of the frequency detector  402  shown in FIG.  13 . The frequency detector of FIG. 14 comprises first and second square calculators  410  and  412 , an adder  414 , an amplifier  416  and a divider  418 . 
     The first square calculator  410  of FIG. 14 squares the signal U 1  output from the second delay  44  (see FIG. 4) and outputs the square result U 1   2  to the adder  414 . The second square calculator  412  squares the signal V 1  output from the fourth delay  48  (see FIG. 4) and outputs the square result V 1   2  to the adder  414 . The adder  414  adds the output of the second square calculator  412  to the output of the first square calculator  410 , and outputs the addition result U 1   2 +V 1   2  to the amplifier  416 . The amplifier  416  amplifies the addition result of the adder  414  and outputs the amplified result to the divider  418 . Here, the divider  418  divides the latched Z signal ZD in the flip-flop  400  by the output of the amplifier  416 , and outputs the divided result as the frequency f CB  of the color burst signal. Here, it can be understood that the latched Z signal ZD corresponds to X i *Y i+1 −X i+1 *Y i  in Equation (10) and the result amplified by the amplifier  416  corresponds to B 2 k f  or            [       A        (   iT   )       2     ]     2          k   f                            
     from Equation (6) or (7). That is, the frequency detector  402  detects the frequency of the color burst signal using the arcsin approximation. Also, in the frequency detector  402  of FIG. 14, the first and second square calculators  410  and  412  may square the outputs U 2  and V 2  of the first and third delays  42  and  46  respectively, instead of the outputs U 1  and V 1  of the second and fourth delays  44  and  48 . 
     The comparator  404  of FIG. 13 compares the frequency f CB  of the color burst signal output from the frequency detector  402  with a predetermined frequency and outputs the comparison result as the check result LD or LD′. For example, the comparator  404  outputs the check result (LD or LD′) as a logic “high” if the frequency of the color burst signal is lower than a predetermined frequency, and as a logic “low” if the frequency of the color burst signal is higher than the predetermined frequency. Here, if the line checking unit of FIG. 13 is the first or second line checking unit  106  or  166 , the predetermined frequency is 4.33 MHz, that is, the average value of the sub-carrier frequency f ob  of the analog CVBS having the first color difference signal, and the sub-carrier frequency f or  of the analog CVBS having the second color difference signal. However, if the line checking unit of FIG. 13 is the third or fourth line checking unit  306  or  346 , the predetermined frequency may be 4.25 MHz which is the sub-carrier frequency f ob  of the analog CVBS having the first color difference signal. 
     FIG. 15 is a circuit diagram of an example of the first gain and offset controller  110  of FIG. 7 or the second gain and offset controller  170  of FIG. 9 according to the present invention. The first gain and offset controller of FIG. 15 comprises a multiplier  422 , an adder  426  and multiplexers  420  and  424 . 
     The multiplexer  420  of FIG. 15 selects one of the first and second gain values D bgain  and D rgain  for the first and second color difference signals in response to the check result LD from the line checking unit, and outputs the selected gain value to the multiplier  422 . The multiplexer  424  selects one of the first and second offset values D boffset  and D roffset  for the first and second color difference signals in response to the check result LD and outputs the selected result to the adder  426 . The multiplier  422  multiplies the signal Z output from the third or fourth frequency differentiator  104  or  164  by the output of the multiplexer  420  and outputs the product to the adder  426 . The adder  426  adds the output of the multiplexer  424  to the product of the multiplier  422  and outputs the addition result as the gain-adjusted and offset-compensated Z signal Z′ to the fourth or sixth de-emphasizer  112  or  172 . Here, the first or second gain and offset controller  110  or  170  of FIG. 15 adjusts the first and second gain values so as to normalize B 2 k f  to 1. 
     FIG. 16 is a circuit diagram of an example of the third gain corrector  310  of FIG. 11 or the fourth gain corrector  350  of FIG.  12 . The gain adjustor of FIG. 16 comprises a multiplexer  430  and a multiplier  432 . 
     In FIG. 16, the multiplexer  430  selects one of the first and second gain values D bgain  and D rgain  for the first and second color difference signals in response to the check result LD′ from the line checking unit, and outputs the selected gain value to the multiplier  432 . The multiplier  432  multiplies the signal Z output from the fifth or sixth frequency differentiator  304  or  344  by the output of the multiplexer  430 , and outputs the product as the gain-adjusted Z signal Z″ to the eighth or tenth de-emphasizer  312  or  352 . 
     Here, the third or fourth gain corrector  310  or  350  of FIG. 16 adjusts the gains values D bgain  and D rgain  so as to normalize B 2 k f  to 1. 
     Next, the structure and operation of the first, second, third or fourth color difference signal reproducer  108 ,  168 ,  308  or  348  according to an embodiment of the present invention will be described. 
     FIG. 17 is a circuit diagram of the first, second, third or fourth color difference signal reproducer  108 ,  168 ,  308  or  348 . The color difference signal reproducer of FIG. 17 comprises a line memory  450  and multiplexers  452  and  454 . 
     In FIG. 17, the line memory  450  stores the signal Z for the previous line. That is, the line memory  450  stores the Z signal for the previous line, output from the fourth, sixth, eighth or tenth de-emphasizer  112 ,  172 ,  312  or  352 . Here, the multiplexer  452  selectively outputs one of the Z signal for the current line output from the corresponding de-emphasizer or the Z signal for the previous line stored in the line memory  450 , as the frequency-demodulated first color difference signal D b ′, in response to the check result LD or LD′. Further, the multiplexer  454  selectively outputs the other of the Z signal for the current line, or the Z signal for the previous line stored in the line memory  450  (i.e., the one not selected by the multiplexer  452 ), as the frequency-demodulated second color difference signal D r ′, in response to the check result LD or LD′. For example, if the multiplexer  452  selects the Z signal stored in the line memory  450 , the multiplexer  454  selects the Z signal for the current line. Meanwhile, if the multiplexer  452  selects the Z signal for the current line, then multiplexer  450  selects the previous Z signal stored in the line memory  450 . Here, the first and second color difference signals D b ′ and D r ′ are output in lines. That is, D b ′=D b (0)˜D b (n) and D r ′ Dr(0)˜D r (n). 
     As described above, the frequency demodulation apparatus and the method thereof according to the present invention utilize the arcsin approximation, so the frequency is demodulated without the need for a phase locked loop (PLL) or a look up table, thereby reducing the volume of hardware and the manufacturing costs. Also, the frequency demodulation apparatus and the method thereof according to the present invention can be applied to demodulate the frequency of the color difference signals transmitted according to by the SECAM protocol. 
     While this invention has been particularly shown and described with references to preferred embodiments thereof, it will be understood by those skilled in the art that various changes in form and in details may be made therein without departing from the spirit and scope of the invention as defined by the appended claims.