Abstract:
Orthogonal frequency division multiplexing (OFDM) has become a popular transmission method for high speed wireless radio transmission, due to its potential for low complexity of transmitters and receivers. A method and apparatus are contemplated for cancelling additive sinusoidal disturbances of a known frequency in OFDM receivers which arise e.g. from clock signals that are present for frequency reference, mixer control, and A/D converter control, as well as harmonics and mixing products of those periodic signals, coupling into some point in the receiver chain and appearing as rotating complex exponentials superimposed to complex baseband receive signals. According to the inventive method and apparatus an estimation of an amplitude and phase of a disturbing superimposed tone with a known frequency is obtained and the amplitude and phase estimation is used to cancel the spurious tone preventing a degradation of receiver sensitivity while achieving low implementation complexity.

Description:
[0001]    The present invention relates to a method and an apparatus to cancel additive sinusoidal disturbances of a known frequency in OFDM receivers. 
       BACKGROUND OF THE INVENTION 
       [0002]    Orthogonal frequency division multiplexing (OFDM) has become a popular transmission method for high-speed wireless radio transmission, due to its potential for low complexity of transmitters and receivers, paired with robustness under severe multi-path conditions. A more detailed discussion on OFDM in found in S. B. Weinstein and P. M. Ebert: Data transmission by frequency-division multiplexing using the discrete Fourier transform. IEEE Trans. Communication Technology, COM-19(5):628-634, Oct. 1971. The wired counterpart, known as discrete multi-tone (DMT) employs similar techniques. The transmitter uses an inverse discrete Fourier transform (IDFT) to generate a multi-carrier signal, and the receiver applies the Discrete Fourier Transform (DFT) to demodulate the data. 
         [0003]    Integrated radio receivers need a large gain and a low noise figure to achieve a high sensitivity. Clock signals which are present for frequency reference, mixer control, and A/D converter control, as well as harmonics and mixing products of these periodic signals, may couple into some point in the receiver chain and appear as rotating complex exponentials superimposed to the complex baseband receive signal. If the level of such tones becomes too high, they may degrade the receiver sensitivity. The frequencies of such disturbing tones originating from the RF receiver itself are directly related to the clock frequencies occurring in the receiver. 
         [0004]    As stated above, unwanted tones superimposed to the received signal may reduce the receiver sensitivity. The safest approach to prevent this problem is to directly avoid the occurrence of such tones. Even the coupling mechanism may be known and a re-spin of the receiver design may be able to reduce the coupling. However, in highly integrated receiver systems the effort to achieve this can be quite high, possibly requiring detailed modelling, design modifications and additional verification. 
         [0005]    A general object of the present invention, therefore, is to mitigate such additive disturbing tones in an OFDM baseband receiver, while achieving low implementation complexity. 
       SUMMARY OF THE INVENTION 
       [0006]    According to an aspect of the present invention there is provided a method for cancelling additive sinusoidal disturbances in OFDM receivers as claimed in claim  1 . According to a further aspect of the present invention there is provided an apparatus as claimed in claim  16 . The inventive method and apparatus obtain an estimation of an amplitude and phase of a disturbing superimposed tone, whose frequency is known, and use such amplitude and phase estimation values to cancel the tone such that receiver sensitivity degradation is avoided. 
         [0007]    In accordance with the invention the implementation is made in a way to achieve a low complexity, which translates into low overhead power consumption in applying the method. 
     
    
     
         [0008]    Additional features and advantages of the present invention will be apparent from the following detailed description of specific embodiments which is given by way of example and in which reference will be made to the accompanying drawings, wherein: 
           [0009]      FIG. 1  is a schematic block diagram of an OFDM receiver, in which the present invention may be implemented; 
           [0010]      FIG. 2  is a block diagram of a typical known OFDM demodulator; 
           [0011]      FIG. 3  shows a block diagram of an OFDM baseband receiver according to a preferred embodiment of the present invention; 
           [0012]      FIG. 4  is a block diagram illustrating in more detail a preferred embodiment of the spur cancellation unit of  FIG. 3  according to the present invention; and 
           [0013]      FIG. 5  is a block diagram showing another preferred embodiment of the spur cancellation unit of  FIG. 3  according to the present invention. 
       
    
    
     DETAILED DESCRIPTION 
       [0014]      FIG. 1  shows a schematic block diagram of an OFDM receiver  1  in which the present invention may be implemented. An analog OFDM radio signal is received via an antenna  20  and is fed into a radio receiver  30  where it is converted to a digital complex baseband signal. Typically, radio receiver  30  consists of a low noise amplifier, a mixer which is controlled by a local oscillator, a band selection filter, further amplifier stages and optionally a second mixer, an analog-to-digital converter, and a digital decimation filter. Radio receiver  30  outputs a digital complex baseband signal. This signal is fed into a digital OFDM baseband demodulator  40 , from where demodulated data are output. 
         [0015]      FIG. 2  is a block diagram of a typical OFDM demodulator  40  as shown in  FIG. 1 , as it is known from the prior art. The input signal which is a digital complex baseband signal supplied from radio receiver  30  of  FIG. 1  is fed to a guard interval removal unit  410  where it is cut into blocks of samples of a length corresponding to the OFDM symbol period. Then, the guard period of each such sample block is removed, and a Discrete Fourier Transform (DFT) is performed on each remainder of the sample blocks in a Discrete Fourier Transform unit  420 . DFT unit  420  outputs data comprised of symbols which are received on respective OFDM sub-carriers in a data equalization unit  430 . Optionally, channel estimation for all sub-carriers of interest is performed in a channel estimation unit  425  prior of being fed into data equalization unit  430 . After equalization the sub-carrier symbols are fed from data equalization unit  430  to a symbol demapper  440  which outputs soft bits to be fed to a decoder. Discrete Fourier Transform in DFT unit  420  is typically implemented as a Fast Fourier Transform (FFT). This kind of OFDM demodulator is well known in prior art. However, it has the drawback that it is not robust against sinusoidal disturbances which typically occur by coupling of periodic voltages or currents into the RF signal path. Such sinusoidal disturbances appear at the input of OFDM baseband receiver  40  as superimposed complex rotating exponentials. Depending on the level and frequency of such disturbances, a large number of information symbols may be corrupted. This degrades the sensitivity of the receiver. 
         [0016]      FIG. 3  shows a OFDM baseband receiver  40 A modified according to the invention. OFDM baseband receiver  40 A is similar to OFDM baseband receiver  40  of  FIG. 2  described above except that it additionally includes a spur cancellation unit  500  right behind DFT unit  420 . The function of this spur cancellation unit  500  is to estimate both an amplitude and a phase of a superimposed rotating exponential of known frequency which is present at the input of DFT unit  420 , and to eliminate this disturbance. Such tones of known frequency typically originate from harmonics and potential mixing products of periodic signals occurring in the RF front-end. The frequencies of those signals are in a constant ratio with the frequency of the reference clock which is normally used for both the RF front-end and the digital baseband receiver. 
         [0017]    Before referring to  FIG. 4  which illustrates spur cancellation unit  500  in greater detail, some mathematical basis of the functionality of spur cancellation unit  500  will be delineated, first, for the sake of understanding of the operation thereof, as follows. We make the following definitions: 
         [0018]    f T  is the frequency of the disturbing tone, normalized to the sampling frequency; 
         [0019]    N DFT  is the length of the discrete Fourier transform in samples; 
         [0020]    N Guard  is the length of the guard interval in samples; 
         [0021]    N Sym =N DFT +N Guard  is the number of time-domain samples per OFDM symbol; 
         [0022]    k is the sampling time index; 
         [0023]    y(k)=r(k)+z(k) is the complex baseband receive signal input into OFDM demodulator  40 ; with 
         [0024]    r(k) being the actual receive signal including other disturbances like noise; and 
         [0025]    z(k)=A T ·exp(j2π·f T ·k+φ T ) being the disturbing superimposed complex exponential; with 
         [0026]    f T  being the known frequency, and A T  and φ T  being amplitude and phase, respectively, of the disturbing complex exponential, which are to be estimated. 
         [0000]    Assuming that 0≦f T &lt;1, the periodic spectrum of a digital signal allows to map any possible tone onto this range. 
         [0027]    Transformation of a complex exponential z(k)|0≦k&lt;N DFT  via DFT yields the values 
         [0000]    
       
         
           
             
               
                 Z 
                  
                 
                   ( 
                   n 
                   ) 
                 
               
               = 
               
                 
                   A 
                   T 
                 
                 · 
                 
                   exp 
                    
                   
                     ( 
                     
                       j 
                       · 
                       
                         ϕ 
                         T 
                       
                     
                     ) 
                   
                 
                 · 
                 
                   
                     ∑ 
                     
                       k 
                       = 
                       0 
                     
                     
                       
                         N 
                         DFT 
                       
                       - 
                       1 
                     
                   
                    
                   
                     exp 
                      
                     
                       ( 
                       
                         j 
                          
                         
                             
                         
                          
                         2 
                          
                         
                           π 
                           · 
                           k 
                           · 
                           
                             ( 
                             
                               
                                 f 
                                 T 
                               
                               - 
                               
                                 n 
                                 
                                   N 
                                   DFT 
                                 
                               
                             
                             ) 
                           
                         
                       
                       ) 
                     
                   
                 
               
             
             , 
           
         
       
     
         [0000]    where
 
n denotes the element index in the resulting vector.
 
Rewriting this equation as
 
         [0000]        Z ( n )=[ A   T ·exp( j·φ   T )/ N   DFT   ]·W ( n ) 
         [0000]    splits it into the amplitude/phase factor (A T ·exp(j·φ T )/N DFT ) and the weighting pattern 
         [0000]    
       
         
           
             
               
                 W 
                  
                 
                   ( 
                   n 
                   ) 
                 
               
               = 
               
                 
                   1 
                   
                     N 
                     DFT 
                   
                 
                 · 
                 
                   
                     ∑ 
                     
                       k 
                       = 
                       0 
                     
                     
                       
                         N 
                         DFT 
                       
                       - 
                       1 
                     
                   
                    
                   
                     exp 
                      
                     
                       ( 
                       
                         j 
                          
                         
                             
                         
                          
                         2 
                          
                         
                           π 
                           · 
                           k 
                           · 
                           
                             ( 
                             
                               
                                 f 
                                 T 
                               
                               - 
                               
                                 n 
                                 
                                   N 
                                   DFT 
                                 
                               
                             
                             ) 
                           
                         
                       
                       ) 
                     
                   
                 
               
             
             , 
           
         
       
     
         [0000]    which is only determined by the frequency of the disturbing tone (when treating N DFT  as given). We assume that the frequency of the disturbing tone is known, and we need to estimate the amplitude and the phase of the tone.
 
Now consider the receive signal after DFT, which ideally consists only of a superposition of data symbols disturbed by the channel fading and additive noise. Let y(k)|K·N Sym ≦k&lt;K·N Sym +N DFT  denote the DFT input samples of OFDM symbol number K and
 
         [0028]    Y K (n) denote the associated DFT output vector, with 0≦n&lt;N DFT . 
         [0029]    If no additive complex exponential is present, we assume that the output of all OFDM sub-carriers during reception is a zero-mean random process, i.e., 
         [0000]        E{Y   K ( n )}=0 ∀K,n.    
         [0000]    Furthermore we assume that distinct DFT output symbols are statistically independent, i.e., 
         [0000]        E{Y   K     1   ( n   1 )· Y   K     2   ( n   2 )}=0 ∀K   1   ≠K   2   νn   1   ≠n   2    
         [0030]    Three key ideas are applied for estimation of a superimposed disturbing complex exponential: 
         [0000]    1. The scalar product of the DFT output vector Y K (n) with the pattern W(n|f T ), 
         [0000]    
       
         
           
             
               P 
               K 
             
             = 
             
               
                 ∑ 
                 
                   n 
                   = 
                   0 
                 
                 
                   
                     N 
                     DFT 
                   
                   - 
                   1 
                 
               
                
               
                 
                   
                     Y 
                     K 
                   
                    
                   
                     ( 
                     n 
                     ) 
                   
                 
                 · 
                 
                   
                     W 
                     * 
                   
                    
                   
                     ( 
                     n 
                     ) 
                   
                 
               
             
           
         
       
     
         [0000]    is the projection of the DFT output vector into the direction of the tone and is an estimate of the amplitude and phase factor (A T ·exp(j·φ T )/N DFT ) multiplied with a phase offset term exp(j2π·f T ·K·N Sym ) which is the start of phase of the complex exponential at the beginning of OFDM symbol number K. Hence, with the assumptions made the expectation of the above scalar product is E{P K }=A T ·(j·φ T )/N DFT ·exp(j2π·f T ·K·N Sym ).
 
2. A back-rotation of the scalar product by the start phase yields Q K =P K ·exp(−j2π·f T ·K·N Sym ) which is an estimate of the amplitude and phase with E{Q K }=A T ·(j·φ T )/N DFT  
 
3. Averaging of multiple such back-rotated estimates Q K  reduces the estimation error.
 
         [0031]      FIG. 4  illustrates in more detail a first preferred embodiment of spur cancellation unit of  FIG. 3  according to the invention for cancelling a disturbing complex exponential in an OFDM receiver employing the principles described above. 
         [0032]    An offset phasor F K =exp(j2π·f T ·N Sym ), here with the constant F K =F∀K is input to an offset phasor accumulation unit  510  and is cumulatively multiplied to obtain a sequence of start phasors 
         [0000]    
       
         
           
             
               R 
               K 
             
             = 
             
               
                 
                   ∏ 
                   
                     k 
                     = 
                     
                       - 
                       ∞ 
                     
                   
                   K 
                 
                  
                 
                     
                 
                  
                 F 
               
               = 
               
                 
                   exp 
                    
                   
                     ( 
                     
                       
                         j2π 
                         · 
                         
                           f 
                           T 
                         
                         · 
                         K 
                         · 
                         
                           N 
                           Sym 
                         
                       
                       + 
                       
                         ϑ 
                         0 
                       
                     
                     ) 
                   
                 
                 . 
               
             
           
         
       
     
         [0033]    R K  is fed into a back rotation unit  520  where the complex conjugates of these values are multiplied with the amplitude/phase estimator output values 
         [0000]    
       
         
           
             
               P 
               K 
             
             = 
             
               
                 ∑ 
                 
                   n 
                   = 
                   0 
                 
                 
                   
                     N 
                     DFT 
                   
                   - 
                   1 
                 
               
                
               
                 
                   
                     Y 
                     K 
                   
                    
                   
                     ( 
                     n 
                     ) 
                   
                 
                 · 
                 
                   
                     
                       W 
                       ~ 
                     
                     K 
                     * 
                   
                    
                   
                     ( 
                     n 
                     ) 
                   
                 
               
             
           
         
       
     
         [0000]    from an amplitude and phase estimation unit  570 .
 
Here {tilde over (W)} K (n) is the estimation pattern, which equals W(n) in a first embodiment of the invention, which may however be simplified in another embodiment, as explained below. Furthermore, the estimation pattern may vary from OFDM symbol to OFDM symbol, which is denoted by the index K.
 
         [0034]    The obtained back-rotated amplitude/phase estimates Q K =R K ·P* K  are fed into an Infinite Impulse Response (IIR) linear low-pass filter  530  with a DC gain of one (“History averaging”), controlled by the factors c K  with 0&lt;c K &lt;1. In a first embodiment of the invention, the factors c K  are constant over time, irrespective of K. 
         [0035]    The output values  Q   K  of filter  530  are then supplied to a forward rotation unit  540  and are forward rotated to obtain the estimated amplitude and phase for the current OFDM symbol,  P   K =  Q   K ·  R   K . The filter output value  Q   K  is obtained after a filter memory, because the estimate applied for cancellation in the current OFDM symbol should be based upon only previous OFDM symbols, thus exploiting statistical independency. 
         [0036]    The output value  P   K  of forward rotation unit  540  is then supplied to a pattern weighting unit  550  and is weighted by a cancellation pattern Ŵ K (n), which is, in a first embodiment of the invention, equal to W(n), but which may be simplified in another embodiment, as will be explained below. Further, the cancellation pattern may vary from OFDM symbol to OFDM symbol which is denoted by the index K. Finally, the obtained vector V K (n) from pattern weighting unit  550  is fed into a subtractor  560  and is subtracted from vector Y K (n) output by DFT unit  420  of  FIG. 4  to obtain an output vector after spur cancellation, 
         [0000]        Z   K ( n )= Y   K ( n )− V   K ( n ). 
         [0037]    In another embodiment of the invention the condition E{Y K (n)}=0 may not be satisfied for some pairs (K,n), which is the case if pilot tones are included in the OFDM signal. To prevent the amplitude/phase estimate from becoming biased, the concerned pairs (K,n) shall not be considered in the estimator. 
         [0038]    This is achieved by a modified arrangement shown in  FIG. 5  that illustrates an extension of the spurious tone cancellation arrangement of  FIG. 4 . The components in  FIG. 5  which are the same or equivalent to components of  FIG. 4  described above are designated with the same reference numerals, and a description thereof should not be repeated, for sake of brevity. 
         [0039]    In the arrangement of  FIG. 5 , the output vectors from DFT unit  420  of  FIG. 3  are first fed into a pilot symbol replacement unit  580 . In pilot symbol replacement unit  580  all DFT output values, which shall not be considered in the estimation, are overwritten with the currently available estimate for the respective DFT bin. This is achieved by defining a pattern S K (n) which indicates at what positions the DFT output values shall not be considered for the estimation. The values of this pattern are defined as 0 where the DFT output shall be considered for spur estimation, and otherwise as 1. 
         [0040]    Thus, the functionality of pilot symbol replacement unit  580  may be described by the equation 
         [0000]    
       
         
           
             
               
                 U 
                 K 
               
                
               
                 ( 
                 n 
                 ) 
               
             
             = 
             
               { 
               
                 
                   
                     
                       
                         Y 
                         K 
                       
                        
                       
                         ( 
                         n 
                         ) 
                       
                     
                   
                   
                     
                       
                         
                           ∀ 
                           
                             ( 
                             
                               K 
                               , 
                               n 
                             
                             ) 
                           
                         
                         | 
                         
                           
                             S 
                             K 
                           
                            
                           
                             ( 
                             n 
                             ) 
                           
                         
                       
                       = 
                       0 
                     
                   
                 
                 
                   
                     
                       
                         V 
                         K 
                       
                        
                       
                         ( 
                         n 
                         ) 
                       
                     
                   
                   
                     
                       otherwise 
                       . 
                     
                   
                 
               
             
           
         
       
     
         [0041]    Another embodiment of the invention exploits a degree of freedom in the choice of the pattern W(n), which is to rotate the phase of the entire vector in the complex plane, in order to obtain real-valued coefficients W(n), which reduces the computational complexity. This can also be achieved using the equation 
         [0000]    
       
         
           
             
               W 
                
               
                 ( 
                 n 
                 ) 
               
             
             = 
             
               
                 1 
                 
                   N 
                   DFT 
                 
               
               · 
               
                 
                   ∑ 
                   
                     k 
                     = 
                     0 
                   
                   
                     
                       N 
                       DFT 
                     
                     - 
                     1 
                   
                 
                  
                 
                   
                     exp 
                      
                     
                       ( 
                       
                         j 
                          
                         
                             
                         
                          
                         2 
                          
                         
                           π 
                           · 
                           
                             ( 
                             
                               
                                 
                                   ( 
                                   
                                     k 
                                     - 
                                     
                                       
                                         N 
                                         DFT 
                                       
                                       2 
                                     
                                   
                                   ) 
                                 
                                 · 
                                 
                                   f 
                                   T 
                                 
                               
                               - 
                               
                                 
                                   k 
                                   · 
                                   n 
                                 
                                 
                                   N 
                                   DFT 
                                 
                               
                             
                             ) 
                           
                         
                       
                       ) 
                     
                   
                   . 
                 
               
             
           
         
       
     
         [0042]    In still another embodiment of the invention the complexity of the amplitude/phase estimator  570  is reduced by exploiting the fact that most of the energy of the disturbing rotating exponential of known frequency is concentrated on a few bins at the DFT output. In this embodiment only a subset of DFT output bins, indexed by the set N Est ={n 1 , n 2 , . . . , n N     Est   } ⊂ {0,1, . . . , N DFT −1} 
         [0000]    is used, and the estimation pattern is determined by 
         [0000]    
       
         
           
             
               
                 
                   W 
                   ~ 
                 
                 K 
               
                
               
                 ( 
                 n 
                 ) 
               
             
             = 
             
               { 
               
                 
                   
                     
                       
                         
                           W 
                           K 
                         
                          
                         
                           ( 
                           n 
                           ) 
                         
                       
                       
                         
                           ∑ 
                           
                             n 
                             ∈ 
                             
                               N 
                               Est 
                             
                           
                         
                          
                         
                           
                              
                             
                               
                                 W 
                                 K 
                               
                                
                               
                                 ( 
                                 n 
                                 ) 
                               
                             
                              
                           
                           2 
                         
                       
                     
                   
                   
                     
                       ∀ 
                       
                         n 
                         ∈ 
                         
                           N 
                           Est 
                         
                       
                     
                   
                 
                 
                   
                     0 
                   
                   
                     
                       otherwise 
                       . 
                     
                   
                 
               
             
           
         
       
     
         [0043]    Here, the subscript K indicates that W(n) may vary from OFDM symbol to OFDM symbol. The set N Est  is typically defined such as to collect most of the energy with a limited number of bins, which is achieved by using only the coefficients with the largest absolute values in W(n). In an extreme case, only a single value out of W(n) is used. 
         [0044]    In still another embodiment of the invention the complexity of the pattern weighting/spur subtraction units,  550  and  560 , respectively, is reduced by exploiting the fact that most of the energy of the disturbing rotating exponential of known frequency is concentrated on a few bins at the DFT output, eliminating the need to subtract negligibly small disturbances. In this embodiment, only a subset of DFT output bins indexed by a set 
         [0000]        N   Cancel   ={n   1   , n   2 , . . . ,  n   N     Cancel   } ⊂ {0, 1, . . . ,  N   DFT −1} 
         [0000]    is used, and the cancellation pattern is defined as 
         [0000]    
       
         
           
             
               
                 
                   W 
                   ^ 
                 
                 K 
               
                
               
                 ( 
                 n 
                 ) 
               
             
             = 
             
               { 
               
                 
                   
                     
                       
                         W 
                         K 
                       
                        
                       
                         ( 
                         n 
                         ) 
                       
                     
                   
                   
                     
                       ∀ 
                       
                         n 
                         ∈ 
                         
                           N 
                           Cancel 
                         
                       
                     
                   
                 
                 
                   
                     0 
                   
                   
                     
                       otherwise 
                       . 
                     
                   
                 
               
             
           
         
       
     
         [0045]    Again the subscript K indicates that W(n) may vary from OFDM symbol to OFDM symbol. The set N Cancel  is typically defined to apply to all elements in W(n) where an unacceptable excessive disturbance is expected to occur. In an extreme case, only a single value out of W(n) is addressed. 
         [0046]    In another embodiment of the invention, a fast ring-in of the history averaging low-pass is realized by time-variation of the filter coefficients c K . For example, when the first amplitude/phase estimate is performed at OFDM symbol with K=1, a good choice of a sequence is 
         [0000]    
       
         
           
             
               c 
               K 
             
             = 
             
               { 
               
                 
                   
                     0 
                   
                   
                     
                       K 
                       &lt; 
                       1 
                     
                   
                 
                 
                   
                     
                       1 
                       / 
                       K 
                     
                   
                   
                     
                       1 
                       ≤ 
                       K 
                       &lt; 
                       
                         K 
                         Limit 
                       
                     
                   
                 
                 
                   
                     
                       1 
                       / 
                       
                         K 
                         Limit 
                       
                     
                   
                   
                     
                       K 
                       ≥ 
                       
                         
                           K 
                           Limit 
                         
                         . 
                       
                     
                   
                 
               
             
           
         
       
     
         [0047]    This results in an equal weighting of all incoming samples until the history averaging low-pass has rung in. After ring-in, weighting of filtered samples decays exponentially over time. 
         [0048]    In another embodiment of the invention, each vector of samples subjected to DFT is first cyclically shifted before the DFT is processed, due to the OFDM receiver design. For a cyclic shift by N Shift  samples, the weighting pattern becomes 
         [0000]    
       
         
           
             
               W 
                
               
                 ( 
                 n 
                 ) 
               
             
             = 
             
               
                 1 
                 
                   N 
                   DFT 
                 
               
               · 
               
                 
                   ∑ 
                   
                     k 
                     = 
                     0 
                   
                   
                     
                       N 
                       DFT 
                     
                     - 
                     1 
                   
                 
                  
                 
                   
                     exp 
                      
                     
                       ( 
                       
                         j 
                          
                         
                             
                         
                          
                         2 
                          
                         
                           π 
                           · 
                           
                             ( 
                             
                               
                                 
                                   
                                     
                                       ( 
                                       
                                         
                                           ( 
                                           
                                             k 
                                             - 
                                             
                                               N 
                                               Shift 
                                             
                                           
                                           ) 
                                         
                                          
                                         mod 
                                          
                                         
                                             
                                         
                                          
                                         
                                           N 
                                           DFT 
                                         
                                       
                                       ) 
                                     
                                     · 
                                   
                                 
                               
                               
                                 
                                   
                                     
                                       f 
                                       T 
                                     
                                     - 
                                     
                                       
                                         k 
                                         · 
                                         n 
                                       
                                       
                                         N 
                                         DFT 
                                       
                                     
                                   
                                 
                               
                             
                             ) 
                           
                         
                       
                       ) 
                     
                   
                   . 
                 
               
             
           
         
       
     
         [0049]    All other principles of the invention are applied as described before. 
         [0050]    In another embodiment of the invention the frequency of the disturbing tone changes over time, possibly due to some adaptation of the mixer frequency in the radio front-end. To cope with this, the offset phasor F K  as well as the estimation pattern {tilde over (W)} K (n) and the cancellation pattern Ŵ K (n) are adapted accordingly. 
         [0051]    In another embodiment of the invention, where multiple disturbing sinusoids shall be cancelled, a plurality of spur cancellers, as described above, may be implemented. In this case all amplitude/phase estimations are performed in parallel on the DFT output data, whereas the subtractions of the estimated tones occur sequentially, tone by tone. 
         [0052]    As an example, consider a DVB-H receiver implementation with N DFT =4096, N Guard =1024, N Sym =5120, N Shift =512, with a sampling frequency f sample =48/7 MHz, which is disturbed by a spurious tone at a frequency f Spur =1 MHz. The normalized frequency of the tone is f T =f Spur /f Sample =7/48. The tone frequency corresponds with the OFDM sub-carrier index n T =f T ·N DFT =597⅓. The offset phasor is determined as 
         [0000]    
       
         
           
             
               F 
               K 
             
             = 
             
               
                 exp 
                  
                 
                   ( 
                   
                     j 
                      
                     
                         
                     
                      
                     2 
                      
                     
                       π 
                       · 
                       
                         7 
                         48 
                       
                       · 
                       5120 
                     
                   
                   ) 
                 
               
               = 
               
                 
                   - 
                   
                     1 
                     2 
                   
                 
                 - 
                 
                   j 
                    
                   
                     
                       
                         3 
                       
                       2 
                     
                     . 
                   
                 
               
             
           
         
       
     
         [0000]    The estimation pattern is defined as 
         [0000]    
       
         
           
             
               
                 
                   W 
                   ~ 
                 
                 K 
               
                
               
                 ( 
                 n 
                 ) 
               
             
             = 
             
               { 
               
                 
                   
                     
                       
                         2 
                          
                         π 
                       
                       
                         3 
                          
                         
                           3 
                         
                       
                     
                   
                   
                     
                       
                         for 
                          
                         
                             
                         
                          
                         n 
                       
                       = 
                       597 
                     
                   
                 
                 
                   
                     0 
                   
                   
                     
                       otherwise 
                       , 
                     
                   
                 
               
             
           
         
       
     
         [0000]    the cancellation pattern is defined as 
         [0000]    
       
         
           
             
               
                 
                   W 
                   ^ 
                 
                 K 
               
                
               
                 ( 
                 n 
                 ) 
               
             
             = 
             
               { 
               
                 
                   
                     
                       
                         
                           
                             
                               1 
                               4096 
                             
                             · 
                             
                               
                                 ∑ 
                                 
                                   k 
                                   = 
                                   0 
                                 
                                 4095 
                               
                                
                               
                                 exp 
                                  
                                 
                                   ( 
                                   
                                     j 
                                      
                                     
                                         
                                     
                                      
                                     2 
                                      
                                     
                                       π 
                                       · 
                                       
                                         ( 
                                         
                                           
                                             
                                               
                                                 
                                                   ( 
                                                   
                                                     
                                                       ( 
                                                       
                                                         k 
                                                         - 
                                                         512 
                                                       
                                                       ) 
                                                     
                                                      
                                                     mod 
                                                      
                                                     
                                                         
                                                     
                                                      
                                                     4096 
                                                   
                                                   ) 
                                                 
                                                 · 
                                               
                                             
                                           
                                           
                                             
                                               
                                                 
                                                   f 
                                                   T 
                                                 
                                                 - 
                                                 
                                                   
                                                     k 
                                                     · 
                                                     n 
                                                   
                                                   4096 
                                                 
                                               
                                             
                                           
                                         
                                         ) 
                                       
                                     
                                   
                                   ) 
                                 
                               
                             
                           
                         
                       
                       
                         
                           
                             
                               for 
                                
                               
                                   
                               
                                
                               591 
                             
                             ≤ 
                             n 
                             ≤ 
                             603 
                           
                         
                       
                     
                   
                   
                     
                         
                     
                   
                 
                 
                   
                     0 
                   
                   
                     
                       otherwise 
                       , 
                     
                   
                 
               
             
           
         
       
     
         [0053]    and the minimum filter constant after ring-in is set to 
         [0000]    
       
         
           
             
               C 
               
                 K 
                 , 
                 min 
               
             
             = 
             
               
                 1 
                 10 
               
               . 
             
           
         
       
     
       APPLICATIONS OF THE INVENTION 
       [0054]    The various embodiments of the invention as detailed above may be applied separately or in combination in an OFDM receiver for wireless or wired transmission including, but not limited to, receivers in wireless local area network (WLAN) applications, e.g., according to the IEEE 802.11 standard, in wireless personal area network (WPAN) applications, e.g., according to the IEEE 802.16 standard, in digital TV receivers for, e.g., DVB-T, DVB-H, T-DMB, DMB-T, DAB, in ultra-wideband (UWB) receivers according to the multi-band OFDM alliance (MBOA) standard proposal, etc.