Abstract:
One aspect is a circuit for tuning a resonance frequency of an electrically small antenna and directly exciting the electrically small antenna. The circuit includes a first source configured for providing a constant voltage. The circuit also includes an antenna and a switched capacitor configured for being alternatively alternately coupled to the first source to be charged thereby and to the antenna for exciting the antenna. Another aspect is a transmitter for transmitting a wireless signal using an antenna without using a variable voltage source to excite the antenna. The transmitter includes a first source configured for providing a constant voltage. The transmitter further includes an antenna and a switched capacitor configured for being alternately coupled to the first source to be charged thereby and to the antenna for exciting the antenna and for tuning a resonance frequency of the antenna.

Description:
CROSS REFERENCE TO RELATED APPLICATION 
       [0001]    This application claims the benefit of U.S. Provisional Application No. 61/917,697, entitled “Electro-Mechanical Radio Frequency Transmitter” and filed Dec. 18, 2013, the contents of which application are incorporated herein by reference. 
     
    
     FIELD OF THE INVENTION 
       [0002]    The present invention relates to a transmitter for high-rate data transmission and, more specifically, to a transmitter for transmitting a high-rate data transmission through direct excitation of an antenna. 
       BACKGROUND OF THE INVENTION 
       [0003]    Spark-gap transmitters are known as the oldest transmitters and date from the 1880s. Illustrated in  FIG. 14  is a conventional spark-gap transmitter  1400  comprising an induction coil  1450  having a primary coil and a secondary coil. The transmitter  1400  further comprises a spark gap  1420  connected in parallel with the secondary coil of the induction coil  1450 . The transmitter  1400  further comprises a tuning coil  1460  connected to one or more Leyden jars  1410  in series. The tuning coil  1460  and the one or more Leyden jars  1410  are together connected in parallel with the spark gap  1420 . The secondary coil of the induction coil  1450  and the one or more Leyden jars  1410  together form an LC resonator. The tuning coil  1480  is connected to group  1470  and to an antenna  1480 . 
         [0004]    The tuning circuit  1400  further comprises a telegraph key  1430  and one or more batteries  1440  connected in series. The telegraph key  1430  and the one or more batteries  1440  are together connected in parallel with the primary coil of the induction coil  1450 . The telegraph key  1430  selectively couples and decouples the batteries  1440  from the primary coil of the induction coil  1450  to provide an instantaneous high-voltage pulse to the spark gap  1420 . When a spark takes place across a narrow gap of the spark gap  1420 , the spark energy will be released in the form of heat and electromagnetic radiation to transmit a wireless signal. The signal may be encoded with Morse code. 
         [0005]    Illustrated in  FIG. 15  is a conventional receiver  1500  comprising a head telephone receiver  1510 , a crystal detector  1520 , a variable condenser, and a two slider tuning coil  1560  comprising a coil, a first slider, and a second slider. The coil of the two slider tuning coil  1560  is connected to an antenna  1580 . The first slider is connected to ground  1570 , and the second slider is connected to one port of the variable condenser  1530 . The other port of the variable condenser  1530  is connected to one port of the crystal detector  1520 . The other port of the crystal detector  1520  is connected to the group  1570 . The head telephone receiver  1510  is connected in parallel with the crystal detector  1520 . The receiver  1500  receives the Morse code signal transmitted by the transmitter  1400 . 
       SUMMARY OF THE INVENTION 
       [0006]    In accordance with an aspect of the present invention, there is provided a circuit for tuning a resonance frequency of an electrically small antenna and directly exciting the electrically small antenna. The circuit includes a first source configured for providing a constant voltage. The circuit also includes an antenna and a switched capacitor configured for being alternatively alternately coupled to the first source to be charged thereby and to the antenna for exciting the antenna. 
         [0007]    In accordance with another aspect of the present invention, there is provided a transmitter for transmitting a wireless signal using an antenna without using a variable voltage source to excite the antenna. The transmitter includes a first source configured for providing a constant voltage. The transmitter further includes an antenna and a switched capacitor configured for being alternately coupled to the first source to be charged thereby and to the antenna for exciting the antenna and for tuning a resonance frequency of the antenna. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0008]    For the purpose of illustration, there are shown in the drawings certain embodiments of the present invention. In the drawings, like numerals indicate like elements throughout. It should be understood that the invention is not limited to the precise arrangements, dimensions, and instruments shown. In the drawings: 
           [0009]      FIGS. 1A-1B  illustrate a transmitter configured for transmitting data through direct excitation of an antenna, in accordance with an exemplary embodiment of the present invention; 
           [0010]      FIG. 2  illustrates a circuit model of the transmitter of  FIGS. 1A-1C , the circuit model comprising a switched capacitor, an RLC resonator, a switch, and a source of constant voltage, in accordance with an exemplary embodiment of the present invention; 
           [0011]      FIGS. 3A-3C  respectively illustrate a voltage waveform across the switched capacitor of  FIG. 2 , a voltage waveform across the RLC resonator of  FIG. 2 , and power supplied by the source of constant voltage of  FIG. 2  during initial switching cycles of the switched capacitor when energy stored in the switched capacitor is building up, in accordance with an exemplary embodiment of the present invention; 
           [0012]      FIGS. 4A-4C  respectively illustrate a voltage waveform across the switched capacitor of  FIG. 2 , a voltage waveform across the RLC resonator of  FIG. 2 , and power supplied by the source of constant voltage of  FIG. 2  during switching cycles of the switched capacitor after which energy stored in the switched capacitor has built up, in accordance with an exemplary embodiment of the present invention; 
           [0013]      FIG. 5  illustrates a voltage waveform across the switched capacitor of  FIG. 2  when the constant voltage provided by the source is 1 V, the resistance of the source is 10Ω, and the switch is switched at 50 MHz, in accordance with an exemplary embodiment of the present invention; 
           [0014]      FIG. 6  illustrates a block diagram of a receiver and a transmitter comprising a switched capacitor, an antenna, and a source, in accordance with an exemplary embodiment of the present invention; 
           [0015]      FIGS. 7A and 7B  illustrate an exemplary embodiment of an Electrically-Coupled Loop Antenna (ECLA), in accordance with an exemplary embodiment of the present invention; 
           [0016]      FIG. 8  illustrates a simulation of the transmitter of  FIG. 6 , in accordance with an exemplary embodiment of the present invention; 
           [0017]      FIG. 9A  illustrates a plot of a far-field voltage of the transmitter of  FIG. 6 , as simulated in  FIG. 8 , when the switched capacitor is switched at 25 MHz, in accordance with an exemplary embodiment of the present invention; 
           [0018]      FIG. 9B  illustrates a voltage waveform across the switched capacitor of  FIG. 6 , as simulated in  FIG. 8 , when the switched capacitor is switched at 25 MHz, in accordance with an exemplary embodiment of the present invention; 
           [0019]      FIG. 9C  illustrates power supplied by the source of  FIG. 6 , as simulated in  FIG. 8 , when the switched capacitor is switched at 25 MHz, in accordance with an exemplary embodiment of the present invention; 
           [0020]      FIG. 10A  illustrates a plot of a far-field voltage of the transmitter of  FIG. 6 , as simulated in  FIG. 8 , when the switched capacitor is switched at 50 MHz, in accordance with an exemplary embodiment of the present invention; 
           [0021]      FIG. 10B  illustrates a voltage waveform across the switched capacitor of  FIG. 6 , as simulated in  FIG. 8 , when the switched capacitor is switched at 50 MHz, in accordance with an exemplary embodiment of the present invention; 
           [0022]      FIG. 10C  illustrates power supplied by the source of  FIG. 6 , as simulated in  FIG. 8 , when the switched capacitor is switched at 50 MHz, in accordance with an exemplary embodiment of the present invention; 
           [0023]      FIG. 11A  illustrates a voltage waveform across the switched capacitor of  FIG. 6 , when the transmitter of  FIG. 6  was prototyped using an ECLA as the antenna and the switched capacitor was switched at 2 MHz, in accordance with an exemplary embodiment of the present invention; 
           [0024]      FIG. 11B  illustrates a voltage waveform across the switched capacitor of  FIG. 6 , when the transmitter of  FIG. 6  was prototyped using an ECLA as the antenna and the switched capacitor was switched at 8 MHz, in accordance with an exemplary embodiment of the present invention; 
           [0025]      FIGS. 12A-12F  illustrate measured voltages at a receiving dipole used to measure an electrical field produced by the transmitter of  FIG. 6  when prototyped using an ECLA as the antenna at different switching frequencies, respectively 2 MHz, 4 MHz, 8 MHz. 12 MHz, 20 MHz, and 25 MHz, in accordance with an exemplary embodiment of the present invention; 
           [0026]      FIG. 13A  illustrates a radiated/received voltage in the time domain in a prototype of the transmitter of  FIG. 6  in which the antenna was prototyped as a Planar Inverted-F Antenna (PIFA), in accordance with an exemplary embodiment of the present invention; 
           [0027]      FIG. 13B  illustrates a radiated/received voltage in the frequency domain in a prototype of the transmitter of  FIG. 6  in which the antenna was prototyped as a PIFA, in accordance with an exemplary embodiment of the present invention; 
           [0028]      FIG. 14  illustrates a conventional spark-gap transmitter; and 
           [0029]      FIG. 15  illustrates a conventional receiver used to receive wireless transmissions from the conventional spark-gap transmitter of  FIG. 14 . 
       
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       [0030]    Reference to the drawings illustrating various views of exemplary embodiments of the present invention is now made. In the drawings and the description of the drawings herein, certain terminology is used for convenience only and is not to be taken as limiting the embodiments of the present invention. Furthermore, in the drawings and the description below, like numerals indicate like elements throughout. 
         [0031]    A spark-gap transmitter is advantageous in that it does not require a periodic signal to power its antenna. However, the use of a telegraph key does not provide for high-data-rate transmissions. 
         [0032]    Referring now to  FIGS. 1A through 1C  there is illustrated a circuit  100 , in accordance with an exemplary embodiment of the present invention. The circuit  100  comprises a source  110  of constant voltage, V dc . Connected in series with one another and together in parallel with the source  110  is a resistor  120  having a resistance, R dc , a switch  130 , and a capacitor  140  having a capacitance, C. The circuit  100  further comprises an antenna  150 . The antenna  150  is an electrically small antenna, e.g., an antenna whose ka factor is smaller than one, where k is wave number and a is the radius of the smallest enclosing sphere. In an exemplary embodiment, the switch  130  is a reflective single pole, double throw switch. Thus, the switch  130  comprises a single input and two outputs that are alternately coupled to the input. The switch  130  is chosen to be reflective so that when the input is connected to one of the outputs, the other of the outputs, i.e., the disconnected output, is not terminated so that it reflects any incoming power. 
         [0033]    In  FIG. 1A , the switch  130  is in a first position, designated in  FIG. 1A  as A, in which the antenna  150  is isolated from the capacitor  140  and in which the capacitor  140  is coupled to the resistor  120  and the source  110 . The capacitor  140  is charged by the source  110 . Thus,  FIG. 1A  illustrates the initial charging phase of the capacitor  140 . 
         [0034]    In  FIG. 1B , the switch  130  is in a second position in which the capacitor  140  is isolated from the source  110  and coupled to the antenna  150  instead. Once the maximum electric energy is stored in the capacitor  140 , the switch  130  is switched to the second position, designated in  FIG. 1B  as B, to provide an impulse-like excitation from the capacitor  140  to the antenna  150 . Depending on the Q factor of the antenna  150  and the switching period of the switch  130 , the entire or part of the stored energy within the capacitor  140  will be injected into the near-zone  155  of the antenna  150  (assuming no loss). Simultaneously, the antenna  150  starts to radiate the injected power in the form of an exponentially damped oscillation. Since the capacitor  140  contributes to the tuning mechanism, the frequency, f L , of the power radiated by the antenna  150  is determined by the loading effect of the capacitor  140  on the antenna  150 , as shown in  FIG. 1B . Depending on the time constant of the fields, the radiated power stays above a certain level for a specific amount of time. The criteria for the lower limit of the radiated power may be determined by the required total radiated power. In sum,  FIG. 1B  illustrates the simultaneous application of a pulse of energy to the antenna  150  and tuning of the antenna  150   
         [0035]      FIG. 1C  illustrates the switch  140  again in the first position A in which the capacitor  140  is reconnected to the source  110  to be recharged. During this time, the energy stored in the antenna near-zone  155  provides the radiative power which exponentially decays. However, since the capacitor  140  is not connected to the antenna  150 , the resonant frequency of the antenna  150  changes to f H , which is the same as the original resonant frequency of the antenna  150  when decoupled from the capacitor  140 . 
         [0036]    If the period of switching of the switch  140  is short compared to the time constant of the field  155 , the antenna  150  will radiate continuously while alternating between the two resonant frequencies, f L  and f H , and the radiated power will remain above a certain level. Minimum radiated power level is a function of switching speed. In other words, if the switching rate is high enough, the magnitude of the field  155  does not drop dramatically and the stored energy around the antenna  150  continues to radiate with slight variations in the magnitude. Since the stored energy is already built up in the near-zone, the variation of the resonant frequency at each switching state immediately appears in the far field and therefore the high-Q property of the antenna is not a limiting factor for the radiation bandwidth. 
         [0037]    Referring now to  FIG. 2 , there is illustrated a circuit, generally designated as  200 , that models the circuit  100 , specifically the antenna  150  thereof, using equivalent electronic components, in accordance with an exemplary embodiment of the present invention. The circuit  200  comprises an equivalent circuit  250  for the antenna  150  of the circuit  100 . The equivalent circuit  250  comprises a capacitor  252  having a capacitance, C 1 , an inductor  254  having a inductance, L, and a resistor  256  having a resistance, R L . The equivalent circuit  250  is a high-Q RLC resonator. 
         [0038]    The circuit  200  further comprises the source  110  of constant voltage, V dc , the resistor  120  having a resistance, R dc , the switch  130 , and the capacitor  140  having a capacitance, C 2  (to distinguish such capacitance from the capacitance, C 1 , of the capacitor  252 ). Finally, the circuit  200  comprises a control signal source  210  that provides a switching signal  215  that controls the position of the switch  130 . 
         [0039]    The switch  130  switches the capacitor  140  between the source  110  and the resonator  250 . The resistor  120  is the source impedance and plays an important role in determining the upper limit of the switching rate of the switch  130 . 
         [0040]    The switched capacitor  140  is not only a tuning component but also an intermediate element to collect electric charge from the source  110  during the charging phase and inject it into the resonator  250  during the discharging phase. The total charge accumulated in the switched capacitor  140  is used to excite the resonant frequency of the resonator  250 . The resonant frequency is determined by L and the sum of C 1  and C 2 . Therefore, an efficient power transfer from the source  110  to the load  250  occurs if the switching period is long enough such that the switched capacitor  140  is charged up to a certain maximum voltage level. 
         [0041]    Because the time constant of the charging phase is equal to R dc C 2 , the source impedance, R dc , must be small in order to achieve a high switching rate with an efficient power transfer. The switching moment here is intended to be when the capacitor  140  is sufficiently charged and its voltage is at maximum. In addition, if the voltage across the switched capacitor  140  is at the maximum at the beginning of charging phase. i.e., the moment that the capacitor  140  is connected to the source  110 , the required time to achieve a full charge in the capacitor  140  can be substantially reduced. Thus, the switching rate can be increased. 
       Example 1 
       [0042]    The circuit  200  was simulated using the transient simulator of ADS software. The component values in the circuit  200  were chosen as follows: C 1 =2546 pF, C 2 =1431 pF, L=39.8 pH, R L =50Ω, and R dc =2Ω. These values resulted in two resonant frequencies, 400 MHz and 500 MHz, with Q factors equal to 500 and 400, when the switched capacitor  140  was connected to and disconnected from the RLC circuit  250 , respectively. 
         [0043]    Within the first several cycles, the stored energy in the resonator  250  was built up and afterward, the charging and discharging phases were similarly repeated.  FIGS. 3A-3C  respectively show the voltage waveform across the switched capacitor  140 , the voltage waveform across the load  250 , and the power supplied by the source  110 , which was simulated as a 1 V DC power supply, during initial switching cycles when the energy stored in the capacitor  140  was being built up.  FIGS. 4A-4C  respectively show the voltage waveform across the switched capacitor  140 , the voltage waveform across the load  250 , and the power supplied by the source  110 , which was simulated as a 1 V DC power supply, during switching cycles after the energy stored in the capacitor  140  was built up. 
         [0044]    The switching signal  215  used in the simulation was a 50 MHz periodic pulse with 50% duty cycle. In  FIGS. 3A-3C , the plotted waveforms were the result of initial switching cycles of the switch  130  in which the stored energy in the LC pair (comprising the inductor  254 , the capacitor  252 , and the capacitor  140 ) was being built up.  FIGS. 4A-4C  show the same waveforms when the stored energy was already built up. It can be seen in  FIG. 4B  that the voltage at the load  250  was a frequency modulated waveform whose amplitude was close to 1 V for both frequencies. The voltage across the switched capacitor  140 , as depicted in  FIG. 4A , indicates that if the switching moment coincides with the maximum voltage, the charging time will be shorter than the case of zero voltage and therefore the rate of switching can be increased. 
       End of Example 1 
       [0045]    To study the impact of the source resistor  120  on the switching rate, a case in which the actual time for a full-charge is longer than the duration of the charging phase is considered.  FIG. 5  shows the voltage of the switched capacitor  140  when the voltage of the source  110 , V dc , is 1 V; the resistance, R dc , of the source  110  is 10Ω; and the frequency of the switching signal  215  is 50 MHz. The charging curve associated with the voltage of the switched capacitor  140  can be expressed as 
         [0000]    
       
         
           
             
               
                 
                   
                     
                       V 
                       cha 
                     
                     = 
                     
                       
                         V 
                         
                           d 
                            
                           
                               
                           
                            
                           c 
                         
                       
                       - 
                       
                         
                           ( 
                           
                             
                               V 
                               
                                 d 
                                  
                                 
                                     
                                 
                                  
                                 c 
                               
                             
                             - 
                             
                               V 
                               R 
                             
                           
                           ) 
                         
                          
                         
                            
                           
                             - 
                             
                               
                                 t 
                                 ′ 
                               
                               
                                 τ 
                                 cha 
                               
                             
                           
                         
                       
                     
                   
                   , 
                 
               
               
                 
                   ( 
                   1.1 
                   ) 
                 
               
             
           
         
       
     
         [0000]    where V R  is the voltage of the switched capacitor  140  at the beginning of its charging phase, and t′ is the delayed time originated at an arbitrary starting point of the charging phase. τ cha  is the charging time constant and is equal to τ cha =R dc C 2 . The envelope of exponentially decaying oscillations during the previous discharge phase can be represented by 
         [0000]    
       
         
           
             
               
                 
                   
                     
                       V 
                       discha 
                     
                     = 
                     
                       
                         V 
                         max 
                       
                        
                       
                          
                         
                           - 
                           
                             
                               
                                 t 
                                 ′ 
                               
                               + 
                               
                                 T 
                                 b 
                               
                             
                             
                               τ 
                               discha 
                             
                           
                         
                       
                     
                   
                   , 
                 
               
               
                 
                   ( 
                   1.2 
                   ) 
                 
               
             
           
         
       
     
         [0000]    where V max  is the maximum voltage during the charging phase, and T b  is the bit period which is equal to a half-pulse or charging/discharging duration. 
         [0046]    The τ discha  is the discharging time constant and is equal to τ discha =R L (C 1 +C 2 ). Equating (1.1) and (1.2) at t′=0 results in 
         [0000]    
       
         
           
             
               
                 
                   
                     V 
                     R 
                   
                   = 
                   
                     
                       V 
                       max 
                     
                      
                     
                       
                          
                         
                           - 
                           
                             
                               T 
                               b 
                             
                             
                               τ 
                               discha 
                             
                           
                         
                       
                       . 
                     
                   
                 
               
               
                 
                   ( 
                   1.3 
                   ) 
                 
               
             
           
         
       
     
         [0047]    Therefore, t 0 , the required time for the switched capacitor voltage to rise from V R  to V max  can be found by substituting (1.3) into (1.1) as 
         [0000]    
       
         
           
             
               
                 
                   
                     V 
                     max 
                   
                   = 
                   
                     
                       V 
                       
                         d 
                          
                         
                             
                         
                          
                         c 
                       
                     
                     - 
                     
                       
                         ( 
                         
                           
                             V 
                             
                               d 
                                
                               
                                   
                               
                                
                               c 
                             
                           
                           - 
                           
                             
                               V 
                               max 
                             
                              
                             
                                
                               
                                 - 
                                 
                                   
                                     T 
                                     b 
                                   
                                   
                                     τ 
                                     discha 
                                   
                                 
                               
                             
                           
                         
                         ) 
                       
                        
                       
                          
                         
                           - 
                           
                             
                               t 
                               0 
                             
                             
                               τ 
                               cha 
                             
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   1.4 
                   ) 
                 
               
             
             
               
                 
                   
                     t 
                     0 
                   
                   = 
                   
                     
                       R 
                       
                         d 
                          
                         
                             
                         
                          
                         c 
                       
                     
                      
                     
                       
                         C 
                         2 
                       
                       · 
                       
                         
                           ln 
                           ( 
                           
                             
                               
                                 V 
                                 
                                   
                                     d 
                                      
                                     
                                         
                                     
                                      
                                     c 
                                   
                                    
                                   
                                       
                                   
                                   - 
                                   
                                     V 
                                     max 
                                   
                                 
                               
                                
                               
                                  
                                 
                                   - 
                                   
                                     
                                       T 
                                       b 
                                     
                                     
                                       
                                         R 
                                         L 
                                       
                                        
                                       
                                         ( 
                                         
                                           
                                             C 
                                             1 
                                           
                                           + 
                                           
                                             C 
                                             2 
                                           
                                         
                                         ) 
                                       
                                     
                                   
                                 
                               
                             
                             
                               
                                 V 
                                 
                                   d 
                                    
                                   
                                       
                                   
                                    
                                   c 
                                 
                               
                               - 
                               
                                 V 
                                 max 
                               
                             
                           
                           ) 
                         
                         . 
                       
                     
                   
                 
               
               
                 
                   ( 
                   1.5 
                   ) 
                 
               
             
           
         
       
     
         [0048]    Equation (1.5) gives the condition which in the bit-rate is sufficiently long such that the voltage of the switched capacitor  140  can reach from V R  to V max . 
         [0049]    Because T b ≧t 0 , equation (1.5) can be rewritten as 
         [0000]    
       
         
           
             
               
                 
                   
                     
                       T 
                       b 
                     
                     
                       ln 
                       ( 
                       
                         
                           
                             V 
                             
                               d 
                                
                               
                                   
                               
                                
                               c 
                             
                           
                           - 
                           
                             
                               V 
                               max 
                             
                              
                             
                                
                               
                                 - 
                                 
                                   
                                     T 
                                     b 
                                   
                                   
                                     
                                       R 
                                       L 
                                     
                                      
                                     
                                       ( 
                                       
                                         
                                           C 
                                           1 
                                         
                                         + 
                                         
                                           C 
                                           2 
                                         
                                       
                                       ) 
                                     
                                   
                                 
                               
                             
                           
                         
                         
                           
                             V 
                             
                               d 
                                
                               
                                   
                               
                                
                               c 
                             
                           
                           - 
                           
                             V 
                             max 
                           
                         
                       
                       ) 
                     
                   
                   ≥ 
                   
                     
                       R 
                       
                         d 
                          
                         
                             
                         
                          
                         c 
                       
                     
                      
                     
                       
                         C 
                         2 
                       
                       . 
                     
                   
                 
               
               
                 
                   ( 
                   1.6 
                   ) 
                 
               
             
           
         
       
     
         [0050]    Assuming V max =kV dc , sine 
         [0000]    
       
         
           
             
               
                 T 
                 b 
               
               = 
               
                 1 
                 
                   2 
                    
                   
                       
                   
                    
                   
                     f 
                     s 
                   
                 
               
             
             , 
           
         
       
     
         [0000]    then (1.6) can be rewritten in terms of switching frequency as 
         [0000]    
       
         
           
             
               
                 
                   
                     
                       f 
                       s 
                     
                     · 
                     
                       ln 
                       ( 
                       
                         
                           1 
                           - 
                           
                             k 
                              
                             
                                 
                             
                              
                             
                                
                               
                                 - 
                                 
                                   1 
                                   
                                     2 
                                      
                                     
                                         
                                     
                                      
                                     
                                       f 
                                       s 
                                     
                                      
                                     
                                       
                                         R 
                                         L 
                                       
                                        
                                       
                                         ( 
                                         
                                           
                                             C 
                                             1 
                                           
                                           + 
                                           
                                             C 
                                             2 
                                           
                                         
                                         ) 
                                       
                                     
                                   
                                 
                               
                             
                           
                         
                         
                           1 
                           - 
                           k 
                         
                       
                       ) 
                     
                   
                   ≤ 
                   
                     
                       1 
                       
                         2 
                          
                         
                             
                         
                          
                         
                           R 
                           
                             d 
                              
                             
                                 
                             
                              
                             c 
                           
                         
                          
                         
                           C 
                           2 
                         
                       
                     
                     . 
                   
                 
               
               
                 
                   ( 
                   1.7 
                   ) 
                 
               
             
           
         
       
     
         [0051]    Referring now to  FIG. 6 , there is illustrated a block diagram of a transmitter  600  and a receiver  690 , in accordance with an exemplary embodiment of the present invention. The transmitter  600  comprises a source  610  of constant or generally constant voltage. The transmitter  600  further comprises a switch  630  connected to an input port  652  of the antenna  650 . The transmitter  600  also comprises a capacitor  640  connected to the switch  630  and a data source  620  providing a control signal  625  to the switch  630 . The source  610  also is connected to the switch  630 . The voltage source  610 , data source  620 , control signal  625 , switch  630 , capacitor  640 , and antenna  650  correspond, respectively, to the voltage source  110 , data source  210 , control signal  215 , switch  130 , capacitor  140 , and antenna  150  described above. Thus, the transmitter  600  and its constituent components behave in the aspects in which the circuits  100  and  200  and their constituents behave, as described above. 
         [0052]    The electrostatic stored energy inside the capacitor  640  is used to energize the antenna  650  and send out a pulse when the switch  630  couples the capacitor  640  to the input port  652  of the antenna  650 . The stored energy which is provided to the capacitor  640  by the power supply  610  when the switch  630  couples the power supply  610  to the capacitor  640  converts to a radiating energy in the form of damped resonating electromagnetic fields. The capacitor  640  is charged by the power supply  610  over a certain amount of time and then the stored energy is injected into the antenna  650  by the switch  630 . However, if the antenna  650  is high-Q, a high amount of stored energy within the capacitor  640  will be stored again in the near-zone of the antenna  650  and a portion of it radiates. The near-field stored energy will keep radiating in an exponentially-decaying pattern while the antenna  650  is disconnected from the capacitor  640 . The damping factor of the fields is inversely related to the Q factor of the antenna  650 . Hence, if the antenna has a high Q, a small decay in the radiating power occurs during the time that the capacitor  640  is recharged by the power supply  610 . The radiation mechanism of the transmitter  600  is illustrated in  FIGS. 1A-1C  and described with reference to these figures. 
         [0053]      FIGS. 7A and 7B  illustrate an exemplary embodiment of the antenna  150 ,  650 , generally designated in  FIGS. 7A and 7B  as  700 , in accordance with an exemplary embodiment of the present invention. The antenna  700  is an Electrically-Coupled Loop Antenna (ECLA). 
         [0054]    The antenna  700  is formed from a loop conductor  710 , having dimensions L×L×W. The loop conductor  710  comprises an input port  720  having a height, h. The input port  720  is formed between first and second lower arms  721 ,  722  of the loop  710  that overlap over a length, w c , of the second lower arm  721 . The second arm  722  forms a capacitive plate disposed over the first arm  721 . The input port  720  is connected to the switch  130  in the circuit  100  or to the switch  630  in the transmitter  600 . The antenna  700  is tuned and powered by loading the input port  720  with a charged or partially charged capacitor. 
       Example 2 
       [0055]    The transmitter  600  having an ECLA  700  as the antenna  650  was simulated in CST Microwave Studio and the scattering parameters were taken into Agilent ADS for transient simulations. The measuring probe, a dipole, was located 1 meter away from the simulated antenna in the E-plane to measure the electric field. The measuring dipole was aligned with the co-pol direction and terminated by a 100 KΩ resistor. 
         [0056]      FIG. 8  shows the ADS circuit simulation set-up which used an ideal single pole-double throw (SPDT) switch to switch a 3 pF capacitor between the antenna and a 1 V DC power source with a 2Ω resistance.  FIGS. 9A-9C  show the received far field voltage, the voltage across the switched capacitor, and the power supplied by the DC source when the switching capacitor was switched at a switching frequency of 25 MHz.  FIGS. 10A-10C  show the far field, the voltage across the switched capacitor, and the power supplied by the DC source for a switching frequency of 50 MHz. It can be seen that the far field for both cases was an FSK modulated signal with the same rate as the switching. The carrier frequencies were around 458 MHz and 648 MHz. 
         [0057]    The switched capacitor  640  voltage quickly approached the DC level in the charging phase and resonated during the discharging phase which indicates that it contributed to the resonant frequency of radiated fields. Since the switched capacitor  640  was small, the time constant of the charging capacitor  640  was very short (2×3 pF=6 ps) and hence, the DC resistance was not a limiting factor in this case. Instead, the Q factor of the antenna  650  was desirably high enough such that, during the charging phase of the switched capacitor  640 , the far field benefitted from a small damping factor and the amount of power decay decreased. 
         [0058]    In practical cases, a lower Q antenna is desirably switched at a higher rate to prevent the far field falling off to low levels. Therefore, higher Q will be a desirable design parameter which results in maintaining an almost consistent power in the far field. A very high-rate FSK modulation with desired frequencies is realized by a small antenna and a DC power source. The consumed power, as depicted in  FIGS. 9C and 10C , shows that, due to the current spikes which occur once per a switching cycle, an impulse-like power is transferred from the source  610  to the antenna  650  at the beginning of every charging phase, provided that a high-Q switched capacitor is used. Nevertheless, a continuous radiation is achieved by using the stored energy within the near-field (during the charging phase) and the stored energy within the switched capacitor  640  (during the discharging phase) and simultaneously the resonant frequency can be tuned according to the switched capacitance. 
       Example 3 
       [0059]    The transmitter  600  was prototyped to test its performance. An ECLA was prototyped according to  FIGS. 7A and 7B  with dimensions L=5 cm, W=1.5 cm, w c =1.5 cm, and h=0.79 mm for use as the antenna  650 . Switching circuitry, supported by a Rogers RT/Duroid 5870 substrate with thickness 31 mils and dielectric constant 2.33, was also prototyped for use in the transmitter  600 . An SPDT switch, a HMC194MS8 from Hittite Microwave Corp. which is a reflective switch (i.e. open circuit when off, versus absorptive switches which are terminated by a matched load when off) with On/Off time about 24 ns (which supports a switching frequency up to 40 MHz), was used as the switch  630 . It should be noted that it is desirable to use a reflective switch as the switch  630  in the transmitter  600  because, during the discharging phase, the power source, e.g., a battery, is open circuited and, therefore, no power is consumed. Also, during the charging phase, the ECLA  650  is open circuited with a higher Q compared to the case when it is terminated by a 50Ω load. 
         [0060]    A 3 V DC power source was used as the power source  610 . A 10 pF capacitor was chosen for the capacitor  640 . It was switched between the 3 V DC power source  610  and the input port  720  of the ECLA  650 . 
         [0061]    The voltage across the switched capacitor  640  is shown in  FIGS. 11A and 11B  for two switching frequencies, 2 MHz and 8 MHz, respectively. As expected, the capacitor  640  stored the electric energy by collecting electric charges during the charging phase and transferred the energy to the ECLA  650  during the discharging phase and at the same time contributed to the resonance of the antenna  650  and caused a continuous FSK signal whose rate was a function of switching frequency rather than the antenna  650  bandwidth. Two resonant frequencies were measured, about 140 MHz and 205 MHz for the loaded and unloaded antenna  650 , respectively. 
         [0062]      FIGS. 12A-12F  show the measured voltage at a receiving dipole at different switching frequencies, respectively 2 MHz, 4 MHz, 8 MHz, 12 MHz, 20 MHz, and 25 MHz. As illustrated in the measurement results, a switching rate of 25 MHz (50 Mb/s) can be easily obtained and it can be even further increased by using a low-loss switch with improved performance along with a high-Q capacitor. 
       Example 4 
       [0063]    Above, an exemplary embodiment of the antenna  150 ,  650  as an ECLA is described. Other exemplary embodiments of the antenna  150 ,  650  are contemplated. For example, in another exemplary embodiment, the antenna  150 ,  650  is a Planar Inverted-F Antenna (PIFA). 
         [0064]    A simulation of the transmitter  600  having a PIFA as the antenna  650  was conducted. The capacitor  640  was chosen to be 4 pF. The switch  630  was switched at a rate of 50 MHz. The power supply  610  was a 1 V DC source. The self-resonance frequency of the PIFA  650  was 600 MHz. When loaded by the 4 pF capacitor, the resonance frequency of the PIFA  650  changed to 400 MHz. The Q factors for the antenna  650  were 63 at the 600 MHz resonance frequency and 112 at the 400 MHz resonance frequency. A smooth FSK signal was obtained at the receiving side, as illustrated in  FIGS. 13A and 13B , which illustrate the radiated/received voltage in the time domain and frequency domain, respectively. 
       CONTEMPLATED USES AND CONCLUSION 
       [0065]    The exemplary embodiments and examples described herein demonstrate a minimized architecture for high-rate transmission through a small antenna. It is shown that since a transient radiation can be achieved by an initial excitation of the antenna, a DC power supply can be applied to excite the fundamental resonance of the antenna at the input port of the antenna. No voltage controlled oscillator or variable voltage source is needed to excite the antenna. Rather, a switched capacitor is used to transfer the energy from the DC source to the antenna and provide a fast frequency-shift keying (FSK) modulation. This technique directly utilizes the DC power supply to deliver the radiation power with minimum number of components and hence, the overall size of the transmitter is reduced. Furthermore, the source of constant voltage, e.g., a battery, is used only to charge the switched capacitor and therefore has a very short duty cycle if the capacitor is high-Q. 
         [0066]    Embodiments of the transmitter herein are contemplated for use in high-temperature environments, such as in a jet engine or at the tip of a drill bit used for drilling an oil well. In such embodiments, the power source can be replaced with a thermocouple junction. Other contemplated uses include electric turbines and motors or in the automotive industry. The embodiments of the transmitters described herein may be used for any application in which a low-cost transmitter may be used. The transmitters described herein are low cost because they do not use sophisticated RF components. 
         [0067]    These and other advantages of the present invention will be apparent to those skilled in the art from the foregoing specification. Accordingly, it is to be recognized by those skilled in the art that changes or modifications may be made to the above-described embodiments without departing from the broad inventive concepts of the invention. It is to be understood that this invention is not limited to the particular embodiments described herein, but is intended to include all changes and modifications that are within the scope and spirit of the invention.