Abstract:
An electronic circuit combines two or more individual wideband RF receivers or transceiver band circuits to produce a usable instantaneous bandwidth that is wider than the bandwidth of the individual band circuits. The electronic circuit overcomes the difficulties of combining bands to provide low signal distortion across the band edges and throughout the combined instantaneous bandwidth of the two or more individual band circuits. This electronic circuit utilizes an amplitude, time delay, and phase adjustment procedure that uses associated adjustable circuitry to eliminate misalignments between the two or more individual band circuits.

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
     This application claims the benefit of U.S. Provisional Application No. 61/714,398, filed on Oct. 16, 2012 under 35 U.S.C. §119(e), which application is hereby incorporated herein by reference in its entirety. 
    
    
     FIELD OF THE INVENTION 
     This invention relates generally to electronic receivers and transceivers, and, more particularly, to electronic receivers and transceivers that have a plurality of separate band circuits, each operating in separate frequency bands, signals from which are combined together into one contiguous frequency band. 
     BACKGROUND OF THE INVENTION 
     Radio frequency receivers and transceivers are known, including receivers and transceivers that use analog circuits, and receivers and transceivers that use both analog and digital circuits. 
     Some electronic components used in receivers and transceivers present speed limitations that influence circuit architecture. For example, analog-to-digital converters (ADCs) used in receivers and transceivers can have speed limitations that make it impractical or impossible to convert analog signals to digital signals fast enough to allow operation over a sufficiently wide frequency band for all applications. One such application is in electronic warfare (EW) receivers and transceivers, for which operation is desirable over a very wide frequency range, for example, three Gigahertz (GHz) or more. In order to achieve a frequency range (i.e., a frequency band) of three GHz, an ADC must sample at a rate of somewhat more than six GHz, which ADC is presently not available if the ADC simultaneously needs to have high dynamic range and/or small size and weight. 
     When a desired bandwidth exceeds the capability of existing electronic components, one approach uses a circuit architecture that provides and operates in separate bands, each having a narrower bandwidth, and then “stitching” the bands together to form one contiguous frequency band having a wider bandwidth. 
     Stitching of bands presents particular problems for many receivers and transceivers. In particular, separate electronic circuits (i.e., separate band circuits) used to provide the plurality of bands have different characteristics, even if the separate electronic circuits are seemingly identical. For example, particularly at radio frequencies, the separate electronic circuits can generate substantially different amplitudes, substantially different group delays, and substantially different phases. The different characteristics result in undesirable signal distortions upon band stitching, particularly near the intersection of two frequency bands. The distortions can be highly problematic, particularly when a signal, either a narrowband signal or a wideband signal, has one or more frequency components that are at or near the intersection of two adjacent frequency bands. 
     It would be highly desirable to provide a circuit and technique for which frequency bands of separate electronic band circuits can be stitched together, but for which signal distortions due to band stitching are greatly reduced. 
     SUMMARY OF THE INVENTION 
     The present invention provides circuits and techniques for which frequency bands of separate electronics band circuits can be stitched together, and for which signal distortions due to band stitching are greatly reduced. 
     In accordance with one aspect of the present invention, an electronic circuit for receiving and processing a signal includes a first band circuit. The first band circuit includes a first down converter configured to receive a signal representative of the received signal and configured to generate a first down converted signal. The first band circuit also includes a first antialias filter coupled to receive a signal representative of the first down converted signal and configured to generate a first antialiased signal, the first antialias filter having a first upper corner frequency and a first lower corner frequency, a span between which is indicative of a first frequency band. The first band circuit also includes a digital low pass filter coupled to receive a signal representative of the first antialiased signal and configured to generate a low pass filtered signal within the first frequency band. The electronic circuit further includes a second band circuit. The second band circuit includes a second down converter configured to receive a signal representative of the received signal and configured to generate a second down converted signal. The second band circuit further includes a second antialias filter coupled to receive a signal representative of the second down converted signal and configured to generate a second antialiased signal, the second antialias filter having a second upper corner frequency and a second lower corner frequency, a span between which is indicative of a second different frequency band. The second band circuit further includes a digital high pass filter coupled to receive a signal representative of the second antialiased signal and configured to generate a high pass filtered signal within the second frequency band. The digital low pass filter comprises a first corner frequency proximate to the first upper corner frequency of the first antialias filter, and the digital high pass filter comprises a second corner frequency proximate to the second lower corner frequency of the second antialias filter. At least one of the first band circuit or the second band circuit further comprises an up converter coupled to receive a signal representative of the low pass filtered signal or a signal representative of the high pass filtered signal, respectively. The up converter is configured to generate an up converted signal. The electronic circuit further includes a combiner configured to combine a signal within a frequency band representative of the first frequency band with a signal within a frequency band representative of the second frequency band to generate a combined signal within a wider combined frequency band wider than the first frequency band and wider than the second frequency band. The wider combined frequency band has a band stitching region in which the first and second frequency bands overlap or abut. The digital low pass filter and the digital high pass filter have substantially complimentary amplitude and phase responses selected to result in the wider combined frequency band having reduced phase response fluctuations and reduced amplitude response fluctuations in the band stitching region. 
     In accordance with another aspect of the present invention, a method is used in an electronic circuit used for receiving and processing a signal. The method includes, in a first band circuit, down converting a signal representative of the received signal to generate a first down converted signal. The method further includes, in the first band circuit, filtering a signal representative of the first down converted signal with a first antialias filter to generate a first antialiased signal, the first antialias filter having a first upper corner frequency and a first lower corner frequency, a span between which is indicative of a first frequency band. The method further includes, in the first band circuit, low pass filtering, with a digital low pass filter, a signal representative of the first antialiased signal to generate a low pass filtered signal within the first frequency band. The method further includes, in a second band circuit, down converting a signal representative of the received signal to generate a second down converted signal. The method further includes, in the second band circuit, filtering a signal representative of the second down converted signal with a second antialias filter to generate a second antialiased signal, the second antialias filter having a second upper corner frequency and a second lower corner frequency, a span between which is indicative of a second different frequency band. The method further includes, in the second band circuit, high pass filtering, with a digital high pass filter, a signal representative of the second antialiased signal to generate a high pass filtered signal within the second frequency band. The digital low pass filter comprises a first corner frequency proximate to the first upper corner frequency of the first antialias filter. The digital high pass filter comprises a second corner frequency proximate to the second lower corner frequency of the second antialias filter. The method further includes frequency shifting to a higher frequency at least one of the first frequency band or the second frequency band. The method further includes combining a signal within a frequency band representative of the first frequency band with a signal within a frequency band representative of the second frequency band to generate a combined signal within a wider combined frequency band wider than the first frequency band and wider than the second frequency band. The wider combined frequency band has a band stitching region in which the first and second frequency bands overlap or abut. The digital low pass filter and the digital high pass filter have substantially complimentary amplitude and phase responses selected to result in the wider combined frequency band having reduced phase response fluctuations and reduced amplitude response fluctuations in the band stitching region. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The foregoing features of the invention, as well as the invention itself may be more fully understood from the following detailed description of the drawings, in which: 
         FIG. 1  is a block diagram showing a multi-band transceiver for which signals from separate band circuits are stitched together, one band circuit having a digital low pass filter and an adjacent band circuit having a digital high pass filter; 
         FIG. 2  is a block diagram showing a multi-band receiver for which signals from separate band circuits are stitched together, one band circuit having a digital low pass filter and an adjacent band circuit having a digital high pass filter; 
         FIG. 3  is a graph showing an overlap after stitching of two frequency bands of the transceiver of  FIG. 1  or the receiver of  FIG. 2  corresponding to two band circuits, and also showing a high pass filter transfer function and a low pass filter transfer function of the two band circuits after stitching; 
         FIG. 4  is a graph showing the frequency bands and filter transfer functions of  FIG. 3 , but on an expanded frequency scale; 
         FIG. 5  is graph showing a group delay generated by two adjacent band circuits of the transceiver of  FIG. 1  or the receiver of  FIG. 2 , showing antialias filter characteristics within the two band circuits and also showing an ideal combination of the frequency bands of the two band circuits; 
         FIG. 5A  is graph showing a group delay generated by two adjacent band circuits similar to those of the transceiver of  FIG. 1  or the receiver of  FIG. 2 , showing antialias filter characteristics within the two band circuits and also showing a non-ideal combination of the frequency bands of the two band circuits, in which the digital low pass filter and the digital high pass filter of  FIGS. 1 and 2  are omitted; 
         FIG. 6  is a flow chart showing a method of calibrating either the transceiver of  FIG. 1  or the receiver of  FIG. 2 ; and 
         FIG. 7  is a block diagram of an exemplary phase adjustment circuit that can be used within the transceiver and receiver of  FIGS. 1 and 2 . 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Before describing the present invention, some introductory concepts and terminology are explained. As used herein, the term “band circuit” is used to describe one of a plurality of band circuits used together in a common electronic circuit, each of which operates upon the same received radio frequency (RF) signal, but which operate in different frequency bands, i.e., upon different frequency band portions of the received radio frequency signal. The received radio frequency signal can have frequency components that can span, from time to time, or at the same time, frequencies within or throughout more than one of the frequency bands of the individual band circuits. Thus, simply stated, if an input signal has a frequency band of possible frequencies three GHz wide (e.g., centered at about 21 GHz), one band circuit can operate on the lowest one GHz of the input signal, a second band circuit can operate on the middle one GHz of the input signal, and a third band circuit can operate on the highest one GHz of the input signal. Frequency bands of the band circuits can be adjacent, forming, once stitched together, a larger contiguous frequency band upon which the common electronic circuit operates. 
     In contrast, as used herein, the term “channels” is used to describe a narrower frequency partitioning, narrower than a band, which can be performed by individual ones of the band circuits or by a receiver function described below in conjunction with  FIG. 2 . 
     As used herein, the term “complementary” when referring, for example, to amplitude and phase responses of two electronic filters, is used to mean that the amplitude and phase responses of the two electronic filters are designed such that they produce a nearly flat amplitude and group delay response when summed together. This facilitates the goal of band stitching (i.e., combining) signals from two associated band circuits with minimal distortion. In general, this goal is easiest to achieve using a digital low pass filter and a digital high pass filter with equal group delay responses. 
     As used herein, the term “signal” is used to describe an electronic current or voltage (or electromagnetic wave) that changes with time. The signal can be a broadband signal spanning a plurality of frequencies or a narrowband signal spanning one or a small number of frequencies. The signal can be continuous in time of can occur from time to time. The signal can have frequency components from low frequencies, for example, 100 Hertz, to very high frequencies, for example 200 Gigahertz. 
     While examples are given below of radio frequency circuits and techniques, it should be appreciated that similar circuits and techniques can be used at any frequencies, both higher and lower, with similar advantageous results. 
     Referring to  FIG. 1 , an exemplary electronic circuit  100  includes a plurality of band circuits  105 ,  137 ,  151 . A first band circuit  105  is described in detail below and, except for differences described below, can be the same as or similar to other ones of the band circuits  137 ,  151 . 
     The electronic circuit  100  provides a transceiver function. A receiver-only function is described below in conjunction with  FIG. 2 . 
     An exemplary radio frequency circuit  100  for receiving and processing an RF signal  102  includes a first band circuit  105 . The first band circuit  105  includes a first down converter  106  configured to receive a signal  104  representative of the received RF signal  102  and configured to generate a first down converted signal  106   a . The first band circuit  105  also includes a first antialias filter  108  coupled to receive a signal  106   a  representative of the first down converted signal  106   a  and configured to generate a first antialiased signal  108   a . The first antialias filter  108  has a first upper corner frequency and a first lower corner frequency, a span between which is indicative of a first frequency band. The first band channel  105  also includes a digital low pass filter  118  coupled to receive a signal  114   a  representative of the first antialiased signal  108   a  and configured to generate a low pass filtered signal  118   a  within the first frequency band. 
     The exemplary radio frequency circuit  100  also includes a second band circuit  137 . The second band circuit  137  includes a second down converter  138  configured to receive a signal  136  representative of the received RF signal  102  and configured to generate a second down converted signal  138   a . The second band circuit  137  also includes a second antialias filter  140  coupled to receive a signal  138   a  representative of the second down converted signal  138   a  and configured to generate a second antialiased signal  140   a . The second antialias filter  140  has a second upper corner frequency and a second lower corner frequency, a span between which is indicative of a second different frequency band. The second band circuit  137  also includes a digital high pass filter  144  coupled to receive a signal  142   a  representative of the second antialiased signal  140   a  and configured to generate a high pass filtered signal  144   a  within the second frequency band. 
     The digital low pass filter  118  has a first corner frequency proximate to the first upper corner frequency of the first antialias filter  108 , and the digital high pass filter  144  has a second corner frequency proximate to the second lower corner frequency of the second antialias filter  140 . 
     The exemplary radio frequency circuit  100  also includes a combiner  134  (e.g., an RF combiner) configured to combine a signal  132   a  within a frequency band representative of the first frequency band with a signal  149   a  within a frequency band representative of the second frequency band to generate a combined signal  134   a  within a wider combined frequency band. The wider combined frequency band has a band stitching region in which the first and second frequency bands overlap or abut. The band stitching frequency region is described more fully below in conjunction with  FIGS. 3 and 4 . 
     The digital low pass filter  118  and the digital high pass filter  144  have substantially complimentary phase responses selected to result in the wider combined frequency band having reduced phase response fluctuations and reduced amplitude response fluctuations in the band stitching region. 
     In some embodiments, an amount of overlap of the first band (associated with the first antialias filter  108 ) with the second band (associated with the second antialias filter  140 ) is selected to result in the wider combined frequency band having reduced phase response fluctuations and reduced amplitude response fluctuations in the band stitching region. In some embodiments, amount of overlap of the first corner frequency (of the digital low pass filter  118 ) with the second corner frequency (of the digital high pass filter  144 ) is also selected to result in the wider combined frequency band having reduced phase response fluctuations and reduced amplitude response fluctuations in the band stitching region. 
     In some embodiments, the digital low pass filter  118  and the digital high pass filter  144  are finite impulse response (FIR) digital filters. 
     The exemplary radio frequency circuit  100  can also include a radio frequency (RF) transmitter  168  coupled to receive the combined signal  134   a , which can be an RF signal, and configured to generate an RF signal  168   a  that can be provided to an RF antenna (not shown). 
     While the above is described in conjunction with the first and second band circuits  105 ,  137 , respectively, the same description applies to any two band circuits that have adjacent bands. To achieve this end, the digital high pass filter  144  can be part of a digital high pass filter/digital low pass filter combination  144  (i.e., a band pass filter). With this arrangement, the combined filter  144  can form a digital low pass filter that can have a phase complimentary to a phase of a digital high pass filter in a third band, of which a digital high pass filter  158  within an Nth band circuit  151  is representative. Thus, it should be appreciated that any number of band circuits can be stitched together to form a wider combined frequency band. 
     It should also be appreciated that, in some embodiments, the digital low pass filter  118  of the first band circuit  105  and the digital high pass filter  158  of the Nth band circuit  151  can be respective combined digital high pass filter/digital low pass filter combinations the same as or similar to the combined filter  144 . 
     In some embodiments, the electronic circuit  100  can also include a first waveform processor (i.e., a receiver/exciter module  126 ) coupled to receive a signal representative of the low pass filtered signal  118   a  and configured to generate a first transmission signal  126   a  in accordance with the low pass filtered signal  118   a . In some embodiments, the electronic circuit  100  can also include a second waveform processor (i.e., a receiver/exciter module  1145 ) coupled to receive a signal representative of the high pass filtered signal  144   a  and configured to generate a second transmission signal  145   a  in accordance with the high pass filtered signal  144   a . The combiner  134  is coupled to receive a signal  132   a  representative of the first transmission signal  126   a  and a signal  149   a  representative of the second transmission signal  145   a  and configured to generate the combined signal  134   a  in accordance with the signals  132   a ,  149   a.    
     In some embodiments, the electronic circuit  100  can also include a first up converter  132  coupled between the first waveform processor  126  and the combiner  134  to generate a first up converted signal  132   a . In some embodiments, the electronic circuit  100  can also include a second up converter  149  coupled between the second waveform processor  145  and the combiner  134  to generate a second up converted signal  149   a . Thus, the combiner  134  is configured to generate the combined signal  134   a  in accordance with the signal  132   a  representative of the first up converted signal  132   a  and the signal  149   a  representative of the second up converted signal  149   a . It will be understood that the signals  132   a ,  149   a  can be in the same RF frequency band in which the received RF signal  102  resides. 
     Each band circuit  105 ,  137 ,  151  can include a local oscillator (LO) generator  133 ,  151 ,  167 , respectively, coupled to the down converters  106 ,  138 ,  152 , respectively, and to the up converters  132 ,  149 ,  166 , respectively. In some embodiments, each one of the LO generators  133 ,  151 ,  167  generates different frequencies  133   a - 133   b ,  151   a - 151   b ,  167   a - 167   b , respectively. Thus, in some embodiments, the different frequencies of the LO generators  133 ,  151 ,  167  can result in the down converted signals within the different band circuits  105 ,  137 ,  167 , i.e. signals  106   a ,  138   a ,  152   a , being in the same frequency range, and thus, the antialias filters  108 ,  140 ,  154  can be the same. However, due to the different frequencies of the LO generators  133 ,  151 ,  167 , each one of the band circuits  105 ,  137 ,  151  operates on (i.e., passes through) a different portion of the frequency spectrum of the received RF signal  102 . The different portions are stitched back together by operation of the up converters  132 ,  149 ,  166  and the combiner  134  to generate the original frequency spectrum (with modifications) of the received RF signal  102 . 
     In operation, with this arrangement, the waveform processors, e.g.,  126 ,  145 , can reproduce the received RF signal  102  (but at respective down-converted (i.e., IF) frequencies), can alter the reproduced RF signal in desired ways, can up convert the reproduced RF signal, and can generate the combined signal  134   a  having the altered characteristics at the RF frequencies of the received RF signal  102 . In one embodiment, the electronic circuit  100  forms a radar jammer for which a radar signal  102  is received and a modified radar signal  134   a  is transmitted in response thereto. 
     Also, in operation, in order to properly stitch two or more bands of the band circuits  105 ,  137  together, a signal passing through each band circuit should match in amplitude, time delay, and phase so that they can be properly added together, each frequency band adjacent to a next in frequency, at the combiner  134 . 
     The digital low pass filter  118  and the digital high pass filter  144  can be designed to have substantially equal time delays, and corresponding substantially equal phase responses. It will be understood that this time delay and phase relationship can be achieved with the above-described digital low pass and digital high pass FIR filters. However, in other embodiments, the phase relationships can be other than equal, so long as the amplitude and phase responses of the electronic filters are complementary, resulting, as described above, in amplitude and phase responses of the two filters that produce a substantially flat amplitude and group delay response when summed together, and hence facilitate the goal of band stitching of two associated band circuits with minimal distortion. 
     It is known that digital FIR filters have a predetermined time delay, which is equivalent to a linear phase response. This characteristic is useful when attempting to generate complementary amplitude and phase responses of channels that operate upon different frequency bands. Infinite impulse response (IIR) filters (e.g., digital IIR filters) can also be used as long as the resulting high pass and low pass filter responses are complimentary and the design is done carefully. 
     The substantially equal time delay responses result in a well behaved group delay for signal frequencies within the above-described band stitching region in which the bands overlap or abut once stitched. Thus, the stitching of adjacent band circuits and signals within the adjacent band circuits can occur with reduced distortion, compared with circuits that do not have the digital low pass filter  118  and the digital high pass filter  144  with substantially complementary amplitude and phase responses. 
     It should be understood that, for circuits that operate at very high RF frequencies, for example, in a GHz range, even small differences in circuit layout, in circuit capacitance, and in circuit inductance can result in time delay mismatch and resulting phase mismatch between two seemingly identical electronic circuits, even digital circuits. Therefore, as used herein, the term “substantially” when referring, for example, to a “substantially complimentary phase response” refers to phase responses that are equal to within the tolerance limits of real electronic circuits that introduce time and phase delay errors, particularly at RF frequencies. It will be understood that a tolerance band associated with the term “substantially” depends upon the signal frequencies at which the electronic circuits operate and upon the circuit technology used to generate the associated circuits. 
     Taking the entire band circuits  105 ,  137 ,  151 , it will be appreciated that, even for digital circuits with perfect circuit matches and otherwise perfect band stitching, other characteristics of the other electronic modules, e.g., analog circuits, within the different band circuits  105 ,  137 ,  151  can result in error, particularly within the band stitching frequency region, once the bands are stitched together. The different characteristics can include, but are not limited to, different amplitudes in adjacent band circuits, different time delays in the adjacent band circuits, and different phases in the adjacent band circuits. 
     In view of the above, each band circuit can include an amplitude adjustment module, e.g.,  120 , a time delay adjustment module, e.g.,  122 , and a phase adjustment module, e.g.,  124 . These modules can be used in a calibration mode of operation to adjust amplitudes, phases, and time delays among the band circuits. Calibration is described more fully below in conjunction with  FIG. 6 . 
     Each one of the band circuits can also include an analog-to-digital converter, e.g.,  114 , coupled to receive the signal  108   a  from an anti-alias filter, e.g.,  108 , and a digital to analog converter, e.g.,  128 , coupled to receive a processed signal, e.g.,  126   a , from a waveform processor, e.g.,  126 . 
     The digital low pass filter  118 , the amplitude adjustment module  120 , the time delay adjustment module  122 , the phase adjustment module  124 , and the waveform processor  126  are shown to be coupled in series within a digital field programmable gate array (FPGA). However, in other embodiments the identified elements can be within another form of digital integrated circuit, for example, a as a digital signal processor (DSP) or as a custom integrated circuit. In still other embodiments, one or more of the amplitude adjustment module  120 , the time delay adjustment module  122 , or the phase adjustment module  124  can be disposed at different positions within the band circuit  105 . For example, in other embodiments, amplitude adjustments, time delay adjustments, or phase adjustments can be made in the analog domain to the RF input signal  104 , or to the RF output signal  132   a , or at any other similar locations in the RF signal path. In still other embodiments, the amplitude adjustments, the time delay adjustments, or the phase adjustments can be made in the analog domain to the signal  108   a , to the analog-to-digital converter signal  114   a , or to signal  128   a  from the digital-to-analog converter  128 . Essentially, the amplitude adjustments, the time delay adjustments, and/or the phase adjustments can be made any place in the signal chain of each band circuit. 
     It should be understood that the antialias filters, e.g.,  108 ,  140 , by themselves are not well suited to generate the responses needed to sum (i.e., combine) signals from adjacent band circuits. It is quite difficult to design analog bandpass filters, especially rapid rolloff antialias filters, to have the sufficiently complementary relationship between their lowpass and highpass responses required for low distortion bandwidth stitching. Additionally, any analog circuit has significant differences unit-to-unit in production and also undergoes changes in characteristics due to temperature and aging, further distorting the summed frequency response upon band stitching. 
     Referring now to  FIG. 2 , an electronic circuit  200  has elements similar to those of the electronic  100  of  FIG. 1 . However, the electronic circuit  200  provides only a receiver function, without a transmit function. Thus, the electronic circuit  200  provides a down conversion the same as or similar to the down conversion described above in conjunction with  FIG. 1 , but provides a different sort of up conversion that stitches the bands together in the digital IF frequency region rather than in the RF frequency region. 
     An exemplary radio frequency circuit  200  for receiving an RF signal  202  includes a first band circuit  207 . The first band circuit  207  includes a first down converter  208  configured to receive a signal  206  representative of the received RF signal  202  and configured to generate a first down converted signal  208   a . The first band circuit  207  also includes a first antialias filter  210  coupled to receive a signal  208   a  representative of the first down converted signal  208   a  and configured to generate a first antialiased signal  210   a . The first antialias filter  210  has a first upper corner frequency and a first lower corner frequency, a span between which is indicative of a first frequency band. The first band channel  207  also includes a digital low pass filter  222  coupled to receive a signal  218   a  representative of the first antialiased signal  210   a  and configured to generate a low pass filtered signal  222   a  within the first frequency band. 
     The exemplary radio frequency circuit  200  also includes a second band circuit  237  comprising a second down converter  232  configured to receive a signal  230  representative of the received RF signal  202  and configured to generate a second down converted signal  232   a . The second band circuit  237  also includes a second antialias filter  234  coupled to receive a signal  232   a  representative of the second down converted signal  232   a  and configured to generate a second antialiased signal  234   a . The second antialias filter  234  has a second upper corner frequency and a second lower corner frequency, a span between which is indicative of a second different frequency band. The second band circuit  237  also includes a digital high pass filter  240  coupled to receive a signal  236   a  representative of the second antialiased signal  234   a  and configured to generate a high pass filtered signal  240   a  within the second frequency band. 
     The digital low pass filter has  222  a first corner frequency proximate to the first upper corner frequency of the first antialias filter  222  and the digital high pass filter  240  has a second corner frequency proximate to the second lower corner frequency of the second antialias filter  234 . 
     The exemplary radio frequency circuit  200  also includes a combiner  272  configured to combine a signal  230   a  within a frequency band representative of the first frequency band with a signal  244   a  within a frequency band representative of the second frequency band to generate a combined signal  272   a  within a wider combined frequency band having a band stitching region in which the first and second frequency bands overlap or abut. The digital low pass filter  222  and the digital high pass filter  240  have substantially complimentary phase responses selected to result in the wider combined frequency band having reduced phase response fluctuations and reduced amplitude response fluctuations in the band stitching region. 
     In some embodiments, an amount of overlap of the first band (associated with the first antialias filter  210 ) with the second band (associated with the second antialias filter  234 ) is selected to result in the wider combined frequency band having reduced phase response fluctuations and reduced amplitude response fluctuations in the band stitching region. In some embodiments, an amount of overlap of the first corner frequency (of the digital low pass filter  222 ) with the second corner frequency (of the digital high pass filter  240 ) is selected to result in the wider combined frequency band having reduced phase response fluctuations and reduced amplitude response fluctuations in the band stitching region. 
     In some embodiments, the digital low pass filter  222  and the digital high pass filter  240  are finite impulse response (FIR) digital filters. 
     While the above is described in conjunction with the first and second band circuits  207 ,  237 , respectively, the same description applies to any two band circuits that have adjacent bands. To this end, the digital high pass filter  240  can be part of a digital high pass filter/digital low pass filter combination  240  (i.e., a band pass filter). With this arrangement, the combined filter  240  can form a digital low pass filter that can have a phase complimentary to a phase of a digital high pass filter in a third band, of which a digital high pass filter  256  in an Nth band circuit  251 , is representative. Thus, it should be appreciated that any number of band circuits can be stitched together to form a wider band. 
     It should also be appreciated that, in some embodiments, the digital low pass filter  222  of the first band circuit  207  and the digital high pass filter  256  of the Nth band circuit  151  can be respective combined digital high pass filter/digital low pass filter combinations the same as or similar to the combined filter  240 . 
     In some embodiments, the electronic circuit  200  can also include a first up converter  244  coupled between the digital high pass filter  240  and the combiner  224  to generate a first up converted signal  244   a . Thus, the combiner  272  is configured to generate the combined signal  272   a  in accordance with the signal  230   a  representative of the low pass filtered signal  222   a  and the signal  244   a  representative of the first up converted signal  244   a . The signals  230   a ,  244   a  are signals in an IF frequency band below the RF frequency band in which the received RF signal  202  resides. 
     Each band channel  207 ,  237 ,  251  can include a respective local oscillator (LO) generator  212 ,  233 ,  249 , coupled to the down converters  208 ,  232 ,  248 , respectively. In some embodiments, each one of the LO generators  212 ,  233 ,  249  generates different frequencies  212   a ,  233   a ,  249   a , respectively. Thus, in some embodiments, the different frequencies of the LO generators  212 ,  233 ,  249  can result in the different band circuits  207 ,  237 ,  251  operating in the same frequency band, i.e., the antialias filters  210 ,  234 ,  250  can be the same. However, due to the different frequencies of the LO generators  212 ,  233 ,  249 , each one of the band circuits  207 ,  237 ,  251  operates on (i.e., passes through) a different portion of the frequency spectrum of the received RF signal  202 . The different portions are stitched back together by operation of the up converters  244 ,  260  and the combiner  272  to generate the original frequency spectrum of the received RF signal  202  but at an IF frequency band lower in frequency than the frequency band of the received RF signal  202 . 
     The exemplary electronic circuit  200  can also include a receiver  274  coupled to receive the combined signal  272   a  (at IF) and configured to generate a receiver output signal  274   a . The receiver  274  can have selectivity to select one or more channels (having a narrower bandwidth than a band). 
     With this arrangement, the combined signal  272   a  can reproduce the RF received signal  202  (but at respective down-converted (i.e., IF) frequencies), and can generate the combined signal  272   a  representative of the received RF signal  202  but at IF frequencies. In one embodiment, the electronic circuit  200  forms a radar receiver and detector for which a radar signal  202  is received, processed, detected, and classified. The processing, detection, and classification functions are not shown. 
     In operation, as with the circuit  100  of  FIG. 1 , in order to properly stitch two or more bands of the band circuits together, a signal passing through each band circuit should match in amplitude, time delay, and phase so that they can be properly added together, each frequency band adjacent to a next in frequency, at the combiner  272 . 
     The digital low pass filter  222  and the digital high pass filter  240  can be designed to have equal (or substantially equal) time delays, and corresponding equal (or substantially equal) phase responses. It will be understood that this time delay and phase relationship can be achieved with the above-described digital low pass and digital high pass FIR filters. However, in other embodiments, the phase relationships can be other than equal, so long as the amplitude and phase responses of the filters are complementary and hence achieve the goal of band stitching with minimal distortion. 
     It is known that digital FIR filters have a predetermined time delay, which is equivalent to a linear phase response. This characteristic is useful when attempting to generate complementary amplitude and phase responses of channels that operate upon different frequency bands. However, in other embodiments, infinite impulse response (IIR) filters (e.g., digital IIR filters) can be used as so long as the resulting high pass and low pass filter responses are complimentary and the design is done carefully. 
     The substantially equal time delay responses result in a well behaved group delay for signal frequencies within the above-described band stitching region. Thus, the stitching of adjacent band circuits and signals within the adjacent band circuits can occur with reduced distortion, compared with circuits that do not have the digital low pass filter  222  and the digital high pass filter  240  with substantially complementary amplitude and phase responses. 
     For reasons described above in conjunction with  FIG. 1 , each band circuit can include an amplitude adjustment module, e.g.,  224 , a time delay adjustment module, e.g.,  226 , and a phase adjustment module, e.g.,  228 . These modules can be used in a calibration mode of operation to adjust amplitudes, phases, and time delays among the band circuits. Calibration is described more fully below in conjunction with  FIG. 6 . 
     Each one of the band circuits can also include an analog-to-digital converter, e.g.,  218 , coupled to receive a signal, e.g.,  210   a , from an anti-alias filter, e.g.,  210 . 
     The digital low pass filter  222 , the amplitude adjustment module  224 , the time delay adjustment module  226 , the phase adjustment module  228 , and a Hilbert filter  230  are shown to be coupled in series within a digital field programmable gate array (FPGA). However, in other embodiments, the identified elements can be within another form of digital integrated circuit, for example, a custom integrated circuit. In still other embodiments, one or more of the amplitude adjustment module  224 , the time delay adjustment module  226 , and the phase adjustment module  228  can be disposed at different positions within the band circuit  207 . For example, in other embodiments, amplitude adjustments, time delay adjustments, or phase adjustments can be made in the analog domain to the RF input signal  202 . 
     It should be understood that, for reasons described above in conjunction with  FIG. 1 , the antialias filters  210 ,  234 , by themselves, do not provide a good means of matching the phase response of the two band circuits  207 ,  237   
     Referring now to  FIG. 3 , a graph  300  has a horizontal axis with a scale in units of frequency in arbitrary units and a vertical axis with a scale in units of relative power in decibels. Bands shown are indicative of bands at frequency positions representative of the combined signal  134   a  of  FIG. 1  or the combined signal  272   a  of  FIG. 2 . As described above, the bands in the combined signal  134   a  of  FIG. 1  reside at the RF frequencies of the received RF signal  102  of  FIG. 1 , but the bands of the combined signal  272   a  can reside at lower IF frequencies. 
     A first band  302  is representative of the first bands of the first antialias filters  108 ,  210  of  FIGS. 1 and 2 , respectively. The first band  302  has a lower band edge  302   a  and an upper band edge  302   b . A second band  304  is representative of the second bands of the second antialias filters  140 ,  234  of  FIGS. 1 and 2 , respectively. The second band  304  has a lower band edge  304   a  and an upper band edge  304   b.    
     From discussion above, it should be appreciated that, in some embodiments the bands  302 ,  304  can overlay each other before up conversion by the up converters  132 ,  149  of  FIG. 1  and by the up converter  244  of  FIG. 2 . The bands are then separated by the up converters  132 ,  149  of  FIG. 1  and by the up converter  244  of  FIG. 2  and are stitched (i.e., added) together by the combiners  134 ,  272  of  FIGS. 1 and 2 , respectively. The graph  300  is representative of the bands once up converted and stitched together. 
     In some embodiments, the first band  302  and the second band  304  each have bandwidths of about 1 Gigahertz. However, other bandwidths and even unequal bandwidths are possible. In some embodiments of the transceiver electronic circuit  100  of  FIG. 1 , once up converted, the first and second bands  302 ,  304  of the combined signal  134   a  intersect at a stitching frequency region from about 20.0 to about 20.1 Gigahertz. In some embodiments of the receiver electronic circuit  200  of  FIG. 2 , once the second band is up converted, the first and second bands  302 ,  304  of the combined signal  272   a  intersect in a stitching frequency region from about 1.9 to about 2.0 Gigahertz. However, different stitching overlap regions are possible, including stitching overlap regions at different frequencies and with different amounts of overlap, including zero overlap. 
     A curve  306  is representative of a band edge and a corn frequency of the digital high pass filters  144 ,  240  when translated by operation of the up converters  149 ,  244 . It should be apparent that the corner frequencies of the high pass filters  144 ,  240  are proximate in frequency to the lower band edge  304   a  of the second band  304  (i.e., of the second antialias filters  140 ,  234 ). 
     A curve  308  is representative of a band edge and a corner frequency of the digital low pass filter  118  when translated by operation of the up converter  132  and is representative of a band edge and corner frequency of the digital low pass filter  222 , which does not undergo an up conversion. It should be apparent that the corner frequencies of the low pass filters  118 ,  222  are proximate in frequency to the upper band edge  302   b  of the first band  302  (i.e., of the first antialias filters  108 ,  210 ). 
     A band stitching region (i.e., crossover region) is primarily determined by the response  306  of the digital low pass filter (e.g.,  118  of  FIG. 1 ) and the response  308  of the digital high pass filter (e.g.,  144  of  FIG. 1 ). The band stitching region is more fully described below in conjunction with  FIG. 4 . 
     It will be apparent that a signal that occurs in the received RF signals  102 ,  202  of  FIGS. 1 and 2  that has a frequency in or corresponding to the band stitching region (crossover of low pass and high pass filter responses) can undergo distortion unless the first and second above-described band channels are matched in amplitude, phase, and time delay within the band stitching region. 
     Referring now to  FIG. 4 , in which like element so  FIG. 3  are shown having like reference designations, the bands and band edges are again shown, but in expanded form. 
     A definition of a frequency extent of the band stitching region is arbitrary and can be defined in a variety of ways. For example, a width (in frequency) of the band stitching region can be defined as a frequency region between points  402 ,  404 . The point  402  can be representative of a point on the low pass filter response  308  that is one dB below an average in-band response (e.g., zero dB) of the low pass filter response  308 . The point  404  can representative of a point on the high pass filter response  306  that is one dB below an average in-band response of the high pass filter response  306 . However, other points on the filter responses  306 ,  308  can also be used to define the band stitching region. For example, points that are three dB below the average in-band responses can also be used. 
     Referring now to  FIG. 5 , a graph  500  has a horizontal axis with a scale in units of frequency in arbitrary units and a vertical axis with a scale in units of group delay in arbitrary units of time. 
     A curve  502  is representative of a group delay of a circuit similar to the first band circuits  105 ,  207  of  FIGS. 1 and 2 , which occupy the first band  302  of  FIGS. 3 and 4  once up converted. A curve  504  is representative of a group delay of a circuit similar to the second band circuits  137 ,  237  of  FIGS. 1 and 2 , which occupy the second band  304  of  FIGS. 3 and 4  once up converted. However, the curves  502 ,  504  are generated without having the digital low pass filters  118 ,  222  and without having the digital high pass filters  144 ,  240  of  FIGS. 1 and 2 , respectively. It can be seen that, for each band, the group delay increases rapidly beyond the intersection of the two bands (i.e., past the band stitching region). 
     A curve  506  (dark curve) depicts an ideal total group delay of the stitched bands, as is generated after the signal combiner  134  of  FIG. 1  or  272  of  FIG. 2 . The curve  506  is generated assuming the pair of stitched bands has been aligned in amplitude, time delay, and phase. Essentially, the ideal curve  506  is synthesized by “chopping off” the high pass portion of the response of the antialias filters  108 ,  210  ( FIGS. 1 ,  2 , respectively), i.e., the curve  502 , above the band of the digital low pass filters  118 ,  222 , and “chopping off” the low pass portion of the response of the antialias filters  140 ,  234  ( FIGS. 1 ,  2 , respectively), i.e., the curve  504 , below the band of the digital high pass filters (within the high pass/low pass filters  144 ,  240 ). 
     The curve  506  has edge regions  506   a ,  506   b  and a central region  506   c . It can be seen that the group delay at the central region  506   c  (i.e., in the band stitching region) is substantially lower than the group delays of the curves  502 ,  504 . This reduction in group delay in the region  506   c  is indicative of an overlap of the bands of the antialias filters (once up converted), and results in lower signal distortion than if the bands were to abut, as opposed to overlap. 
     It will be understood that the low group delay in the region  506   c  would be accompanied by low fluctuations in an amplitude response of the combined frequency band in the same band stitching region. The low group delay and low amplitude fluctuations correspond to low signal distortion of combined signals in the band stitching region. 
     Referring now to  FIG. 5A , a graph  520  has a horizontal axis with a scale in units of frequency in arbitrary units and a vertical axis with a scale in units of group delay in arbitrary units of time. 
     Curves  522 ,  524  are the same as the curves  502 ,  504  of  FIG. 5 , and are representative of band circuits taken separately and without the above-described digital low pass and digital low pass filters. 
     A curve  526  is representative of a group delay of signals similar to the combined signals  134   a ,  272   a  of  FIGS. 1 and 2 , respectively, and still without the above described complimentary digital low pass filters and digital high pass filters. As indicated, the group delay in a band stitching region  526   c  is very high. It will be understood that the high group delay would be accompanied by high fluctuations in an amplitude response in the same band stitching region. The high group delay and high amplitude fluctuations correspond to high signal distortion of combined signals in the band stitching region. 
     It should be appreciated that  FIG. 6  shows a flowchart corresponding to the below contemplated technique, which would be implemented in the electronic circuits  100 ,  200  of  FIGS. 1 and 2 . Rectangular elements (typified by element  602  in  FIG. 6 ), herein denoted “processing blocks,” represent computer software instructions or groups of instructions. Diamond shaped elements (typified by element  622  in  FIG. 6 ), herein denoted “decision blocks,” represent computer software instructions, or groups of instructions, which affect the execution of the computer software instructions represented by the processing blocks. 
     Alternatively, the processing and decision blocks represent steps performed by functionally equivalent circuits such as a digital signal processor circuit or an application specific integrated circuit (ASIC). The flow diagrams do not depict the syntax of any particular programming language. Rather, the flow diagrams illustrate the functional information one of ordinary skill in the art requires to fabricate circuits or to generate computer software to perform the processing required of the particular apparatus. It should be noted that many routine program elements, such as initialization of loops and variables and the use of temporary variables are not shown. It should be noted that many routine program elements, such as initialization of loops and variables and the use of temporary variables are not shown. It will be appreciated by those of ordinary skill in the art that unless otherwise indicated herein, the particular sequence of blocks described is illustrative only and can be varied without departing from the spirit of the invention. Thus, unless otherwise stated the blocks described below are unordered meaning that, when in possible, the steps can be performed in any convenient or desirable order. 
     Referring now to  FIG. 6 , a method  600  depicts an alignment adjustments performed in individual band circuits that enables a successful combination (i.e., stitching) of adjacent bands. At block  602 , a signal is injected at a band crossover frequency. At blocks  604  and  606 , a signal amplitude is measured in each band circuit. At blocks  608 ,  610 ,  612 , amplitudes in the two band circuits are equalized. At blocks  614  and  616 , a group delay is measured in each band circuit. At blocks  618 ,  620 ,  622 , the group delay in the two band circuits are equalized, which can be accomplished by retarding an integer number of samples in a selected one of the two band circuits. At blocks  624  and  626 , a phase is measured in each band circuit. The phase can be considered to be a fine delay. At blocks  628 ,  630 ,  632  the phase in the two band circuits is equalized, which can be accomplished with a phase shift circuit. An exemplary digital phase shift circuit is described below in conjunction with  FIG. 7 . However, in some embodiments, the phase shift circuit can alternatively be implemented with an analog circuit. 
     In particular, at block  602 , signal is provided to the electronic circuit, for example, the electronic circuit  100  of  FIG. 1  or the electronic circuit  200  of  FIG. 2 . The provided signal can be indicative of the input signal  102  of  FIG. 1  or the input signal  202  of  FIG. 2 . For example, the input signal can be an RF chirp signal spanning the bands of interest. 
     At block  604 , an amplitude of the signal in a first band circuit, for example, within the first band circuit  105  of  FIG. 1 , is measured. 
     At block  606 , an amplitude of the signal in a second band circuit, for example, within the second band circuit  137  of  FIG. 1 , is measured. 
     At block  608 , amplitudes of the signal in the two band circuits are compared. If the amplitude of the signal in the first band circuit is larger than the amplitude of the signal in the second band circuit, then the process continues to block  610 . 
     At block  610 , attenuation is added to the first band circuit, for example, by way of the amplitude adjustment module  120  of  FIG. 1 . Attenuation can be added, for example, in the digital domain by bit shifting or by multiplying. 
     At block  608 , if the amplitude of the signal in the first band circuit is not larger than the amplitude of the signal in the second band circuit, then the process continues to block  612 . 
     At block  612 , attenuation is added to the second band circuit. 
     At block  614 , a group delay of the signal in the first band circuit is measured. 
     At block  616 , a group delay of the signal in the second band circuit is measured. 
     At block  618 , group delays of the signal in the two band circuits are compared. If the group delay of the signal in the first band circuit is larger than the group delay of the signal in the second band circuit, then the process continues to block  620 . 
     At block  620 , a time delay is added to the first band circuit, for example, by way of the time delay adjustment module  122  of  FIG. 1 . It will be understood that a time delay can be added to a channel by introducing a gate delay or a register delay synchronous with a clock signal. 
     At block  618 , if the group delay of the signal in the first band circuit is not larger than the group delay of the signal in the second band circuit, then the process continues to block  622 . 
     At block  622 , a time delay is added to the second band circuit. 
     At block  624 , a phase of the signal in the first band circuit is measured. 
     At block  626 , a phase of the signal in the second band circuit is measured. 
     At block  628 , phases of the signal in the two band circuits are compared. If the phase of the signal in the first band circuit is larger (i.e., has more phase) than the phase of the signal in the second band circuit, then the process continues to block  630 . 
     At block  630 , phase is added to the first band circuit, for example, by way of the phase adjustment module  124  of  FIG. 1 . Phase can be added in a number of ways. An exemplary circuit that can add phase to a band circuit is described below in conjunction with  FIG. 7 . 
     At block  628 , if the phase of the signal in the first band circuit is not larger than the phase of the signal and the second band circuit, then the process continues to block  632 . 
     At block  632 , phases added to the second band circuit. 
     Referring now to  FIG. 7 , an exemplary phase adjustment module  700  can be the same as or similar to any of the above-described phase adjustment modules. The exemplary phase adjustment module  700  is in the form of a digital vector modulator. Note that a vector modulator can be implemented either in digital form or in analog form. 
     The phase adjustment module  700  can include a buffer register  704  coupled to receive a digital data signal  702  and configured to generate buffered data  704   a . An IQ module, for example, an IQ finite impulse response (FIR) digital circuit, can be coupled to receive the buffered data  704   a  and configured to generate an I signal and a Q signal, which are ninety degrees apart from each other, regardless of frequency. 
     A first multiplier  708  can be coupled to receive the I signal  706   a , and a second multiplier  710  can be coupled to receive the Q signal  706   b . The first multiplier  708  can also be coupled to receive a first multiplier signal  724   a . The second multiplier  710  can also be coupled to receive a second multiplier signal  726   a . The first multiplier  708  is configured to generate a multiplied signal  708   a  as a product of the I signal  706   a  and the first multiplier signal  724   a . The second multiplier  710  is configured to generate a multiplied signal  710   a  as a product of the Q signal  706   b  and the second multiplier signal  726   a.    
     A summing module  712  is coupled to receive the first and second multiplied signals  708   a ,  710   a  and configured to provide a summed signal  712   a  as a sum of the two input signals. The resultant summed signal  712  is a phase shifted version of the input signal  704   a , adjusted in accordance with a value of a phase adjust signal  722 . 
     A buffer register  714  can be coupled to receive the summed signal  712   a  and configured to generate another buffered signal  714   a.    
     A multiplexer  716  can be coupled to receive the buffered signal  714   a  and coupled to receive the buffered signal  704   a . By way of a bypass control signal  720 , the multiplexer  716  can select as an output signal  716   a  either the buffered signal  714   a , which includes a phase adjustment, or the buffered signal  704   a , which does not include a phase adjustment. 
     Another buffer register  718  can be coupled to receive the output signal  716   a  and configured to generate yet another buffered signal  718   a.    
     In operation, at the summing module  712 , the phase adjustment module  700  is configured to add together different amounts of the I signal and the Q signal depending upon the multiplier signals (i.e., factors)  724   a ,  726   a . The multiplier signals  724   a ,  726   a  can be selected from a plurality of multiplier signals stored within a cosine lookup table (LUT)  724  and a sine lookup table  726 . The selection is made by way of a phase adjustment signal  722 . 
     All references cited herein are hereby incorporated herein by reference in their entirety. 
     Having described preferred embodiments, which serve to illustrate various concepts, structures and techniques, which are the subject of this patent, it will now become apparent to those of ordinary skill in the art that other embodiments incorporating these concepts, structures and techniques may be used. Accordingly, it is submitted that that scope of the patent should not be limited to the described embodiments but rather should be limited only by the spirit and scope of the following claims.