Abstract:
A sensor circuit is coupled to a sensing element for determining a property, such as a dielectric constant, of a fuel suitable where the dielectric constant is used in determining a concentration of ethanol in the gasoline/ethanol blended fuel. The circuit includes an excitation voltage signal generator, a synchronization trigger and a processing circuit configured to generate an output signal indicative of the fuel property (dielectric constant). The excitation voltage signal is applied to the sensing element to produce an induced current signal therethrough. The synchronization trigger is configured to generate a trigger signal when the excitation voltage signal crosses zero volts, at which time the real (resistive) component of the induced current signal is zero. The induced signal is therefore wholly representative of the imaginary component attributable to a capacitance of the sensing element in sensing relation with the fuel, which in turn is dependent on the dielectric constant (and thus ethanol concentration) of the fuel blend itself. The processing circuit is configured to sample the induced signal in response to the trigger signal and produce the output signal. The synchronization scheme provides for a simplified circuit arrangement since there is o need to decompose a signal combining real and imaginary components.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
       [0001]    This application claims the benefit of U.S. provisional application Ser. No. 60/890,112 filed Feb. 15, 2007, presently pending, the disclosure of which is hereby incorporated by reference in its entirety. 
       INCORPORATION BY REFERENCE 
       [0002]    This application incorporates by reference in its entirety U.S. application Ser. No. 10/199,651 filed Jul. 19, 2002, now U.S. Pat. No. 6,693,444 B2 entitled “CIRCUIT DESIGN FOR LIQUID PROPERTY SENSOR” issued Feb. 17, 2004 to Lin et al., owned by the common assignee of the present invention. 
     
    
     TECHNICAL FIELD 
       [0003]    The invention relates in general to sensors used to detect properties of a fuel, and more particularly, to a sensor for detecting the complex impedance of a fuel. 
       BACKGROUND OF THE INVENTION 
       [0004]    Properties of gasoline, such as its conductivity or dielectric constant, are often important for operation of a motor vehicle. For example, flexible fuel vehicles are known that are designed to run on gasoline as a fuel or a blend of up to 85% ethanol (E85). Such properties can be used to determine the concentration of ethanol in the gasoline/ethanol blend and can also determine the amount of water mixed in with the fuel. For example, experimental data shows that the fuel dielectric constant is directly proportional to the ethanol concentration but relatively insensitive to water contamination, while fuel conductivity is very sensitive to water concentration Thus, for these applications and others, there is a need for a fuel sensor that precisely measures the complex impedance of the fuel. 
         [0005]    Current sensor designs have problems handling small capacitance measurements, requiring a relatively large sensing element to increase the signal-to-noise ratio. Further, instead of separately measuring resistance and capacitance, the designs measure total impedance, requiring a relatively high frequency in the 10-100 MHz range to reduce the conductivity impact. Two excitation frequencies are then needed to complete the measurement, a low frequency for resistance measurements and a high frequency for capacitance measurements. 
         [0006]    U.S. Pat. No. 6,693,444 entitled CIRCUIT DESIGN FOR LIQUID PROPERTY SENSOR issued to Lin et al. discloses an improvement to the then-prevailing approaches by providing a single frequency circuit design configured to generate magnitude and phase signals corresponding to the complex impedance of the fuel as shown in  FIG. 5 . Lin et al. disclose a circuit design that characterizes the entire complex impedance of a fuel (i.e., its total complex conductivity). That is, Lin et al. generate both a magnitude signal indicative of total conductivity, including both real (i.e., resistive) and imaginary (i.e., capacitive) parts, as well as a phase signal indicative of the phase angle between an excitation signal and an induced current through the sensing element. While this approach is effective for determining both the dielectric constant as needed for determining ethanol concentration, as well as conductivity as needed for determining water content, further processing is needed to decompose the magnitude signal into its real and imaginary components parts (i.e., one would need to look at just the imaginary part of the magnitude to determine dielectric constant). Additionally, as can be seen in  FIG. 5 , the circuit is relatively complex. 
         [0007]    However, there are certain configurations in the art where just an ethanol concentration sensing system is needed or desired. 
       SUMMARY OF THE INVENTION 
       [0008]    An apparatus in accordance with the present invention provides an improved liquid properties sensor, for example, for determining an ethanol concentration of a gasoline/ethanol blended fuel. The apparatus provides a simplified approach relative to the background art by synchronizing a conductivity current sampling time at the point where the excitation voltage equals zero. When the excitation voltage equals zero, the real component of the total complex conductivity is zero. Accordingly, the total complex conductivity reflects only the imaginary component or part, which as known corresponds to the capacitance of the fuel and thus its dielectric constant. This property can be used to determine ethanol concentration. This synchronization approach obviates the need to decompose the total complex conductivity into its respective real and imaginary parts, and further allows for simplified circuitry (as described herein). 
         [0009]    An apparatus in accordance with the invention includes a sensing element in sensing relation with the fuel, a signal generator, a synchronization trigger and a processing circuit. The signal generator is configured to generate an excitation voltage signal of a predetermined frequency. The excitation voltage signal is coupled to the sensing element to thereby produce an induced signal generally indicative of a total complex conductivity of the fuel. The complex conductivity, at various points in time, may generally comprise both real and imaginary component parts. 
         [0010]    However, the synchronization trigger is configured to generate a trigger signal responsive to the excitation voltage signal at a time when the real component of the complex conductivity is zero. In a preferred embodiment, this time is when the excitation voltage signal is zero. Therefore, it is at this point in time when the contribution of the real component to the total conductivity is zero, leaving just the imaginary component. The processing circuit is configured to receive the induced signal and produce the output signal in response to the trigger signal, thus synchronizing the sampling of the induced signal with the zero crossing. The output signal thus produced comprises just the imaginary component of the total complex conductivity. The timing of the sampling eliminates the need to provide a circuit for determining a phase angle, resulting in a simpler circuit. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0011]    The present invention will now be described by way of example, with reference to the accompanying drawings: 
           [0012]      FIG. 1  is a pictorial representation of one placement of the sensor in an automobile. 
           [0013]      FIG. 2  is a schematic and block diagram of an apparatus for sensing according to the invention. 
           [0014]      FIG. 3  shows, in greater detail, an analog switch portion of  FIG. 3 . 
           [0015]      FIG. 4  is a timing diagram showing input, intermediate, and output signals of the inventive apparatus of  FIG. 2 . 
           [0016]      FIG. 5  is a schematic and block diagram of a known circuit design for a sensing system. 
       
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
       [0017]      FIG. 1  shows a sensor apparatus  96 , which includes a sensing element  60  and a control circuit or electronics  98 , incorporated into an engine control system. Specifically, the sensing element  60  of the sensor apparatus  96  may be located in the fuel tank  12  of a vehicle (not shown) so that it is exposed to fuel. The sensing element  60  is preferably located near the fuel pump  16 , which sends fuel to the engine  20  through fuel line  14 . However, the sensing element  60  can be located elsewhere where it contacts fuel or is otherwise in sensing relation to the fuel, such as in the fuel line  14 . In the illustrated embodiment, the sensing element  60  is submerged in the fuel and excited, and a parameter indicative of a property of the fuel, such as its dielectric constant, is calculated from the induced current measured at a predetermined excitation frequency. It should be understood that in the present disclosure, the sensing element  60 , when submerged in the fuel or otherwise in sensing relation with the fuel, appears to the electronics control circuit  98  as a complex load that can be described for simplicity purposes as a parallel combination of a resistor and a capacitor. As described above, the overall, total impedance presented by the sensing element is a complex impedance inasmuch as it is comprised of a real component (resistive) and an imaginary component (capacitive). It should be further understood that impedance and conductivity are interchangeable terms inasmuch as one is the inverse of the other. To determine a property of a fuel, such as its dielectric constant that can be correlated to a concentration of ethanol, the imaginary component is needed. If the impedance (or conductivity) is determined at a time when both real and imaginary components are present, then the quantity will have to be processed (decomposed) to obtain the desired imaginary component. However, in accordance with the invention, if the sampling is synchronized with the zero crossing of the excitation voltage signal, then the real component of the total complex impedance (or conductivity) will be zero. In this case, there is no need for further processing since only the imaginary component will be present. 
         [0018]    With continued reference to  FIG. 1 , in the illustrated embodiment, the control circuit  98  of the sensor apparatus  96  is configured to excite the sensing element  60  through a shielded cable  22 , such as a coaxial cable, and receives an induced signal from the sensing element  60 . At desired times (e.g., at the time of a zero crossing of the excitation voltage signal), the induced signal is sampled and processed. The control circuit  98  may be configured to include a standard microcontroller, like an engine controller  18  used in automotive applications does, and which includes random access memory (RAM), read-only memory (ROM), input and output means and a processor. The control circuit  98  may be configured to then calculate a capacitance value and supply this value to a diagnostic device or to the engine controller  18 . Alternately, the control circuit  98  may provide its output signal (i.e., corresponding to the capacitance and hence to the dielectric constant of the fuel) to the engine controller  18 , which itself is configured to perform the desired calculations. In either case, the engine controller  18  can manipulate or otherwise employ the output signal, and the dielectric constant of the fuel that may be derived, to control the amount of fuel the engine  20  receives from the fuel tank  12  through the fuel line  14  relative to the intake of air for the operation of an engine  20 . 
         [0019]      FIGS. 2 through 4  illustrate a preferred embodiment configured to produce an output signal indicative of a property of a fuel, such as its dielectric constant, that may be derived from its measured capacitance. The invention may find particular applicability in flex fuel systems for motor vehicles, where, for example, gasoline/ethanol blends of up to 85% (E85) may be expected. 
         [0020]      FIG. 2  is a schematic and block diagram of the apparatus  96 . The apparatus  96  includes a signal generator  100  configured to produce an excitation signal  102  on a node  104 , a sensing element  60  which will produce an induced signal  106  through a node  108 , a synchronization trigger, such as a zero-crossing detector  110 , configured to produce a trigger signal  112  on a node  114 , and a processing circuit  116  configured to produce an output signal  117  destined for receipt by a main controller  18 . The output signal  117  is indicative of a property of a liquid such as its dielectric constant. When the liquid is an ethanol blend fuel, the signal  117  corresponds to a concentration of ethanol in the gasoline/ethanol fuel blend. 
         [0021]    With continued reference to  FIG. 1 , the signal generator  100  is configured to generate the excitation voltage signal  102  at a predetermined frequency. The signal generator  100  includes a sinusoidal source  118  configured to supply a sinusoidal voltage at the predetermined frequency and a buffer  120  having an input coupled to the output of the source  118 . The excitation voltage signal  102  is produced at the same frequency as the AC sinusoid voltage input to the buffer  120 . The predetermined frequency for the excitation voltage signal  102  is in the range of between about 50 kHz and 1 MHz, and more preferably within the range of between about 200 kHz and 300 kHz. As shown in  FIG. 2 , the excitation voltage signal  102  is provided to both the sensing element  60  and the zero-crossing detector  110 . 
         [0022]    The sensing element  60  includes a pair of spaced electrodes  60   a  and  60   b . The electrodes  60   a  and  60   b  may comprise electrically-conductive material, such as various metals known in the art for such purpose. Typical embodiments for electrodes  60   a  and  60   b  may comprise copper-based alloys (e.g., brass). 
         [0023]    The synchronization trigger  110  may comprise a voltage comparator configured as a zero-crossing detector  110 , which is coupled to receive the excitation voltage signal  102 . The zero-crossing detector  110  is configured to generate the trigger signal  112  when the real component of the induced signal is zero (e.g., zero-crossing of the excitation signal), leaving only the imaginary component. In a preferred embodiment, the zero-crossing detector  110  determines this particular point in time as when the excitation voltage signal  102  is within a predetermined trigger range of zero volts, preferably at zero volts. In a constructed embodiment, the range is preferably a mean value (e.g., zero volts). Thus, the zero-crossing detector  110  is configured to output a HIGH digital signal level when the input AC excitation voltage signal  102  is above such mean, predetermined value. The zero-crossing detector  110  is further configured to output a LOW digital signal level when the excitation voltage signal  102  reaches or goes below the mean value (e.g., of zero volts). As shown in  FIG. 4 , the trigger signal  112  transitions high-to-low when the excitation signal  102  transitions high-to-low through zero. 
         [0024]    The processing circuit  116  receives the induced current signal  106  by way of a connection at the node  108  and is configured to produce the output signal  117  synchronized with the trigger signal  112 . The output signal  117  comprises the imaginary component of the total complex conductivity (impedance) observed or measured by the electronics  98  using the sensing element  60 . The output signal  117  may be used by the microcontroller  18  to calculate the fuel capacitance in the first instance, and then to determine indirectly the dielectric constant of the fuel, which is indicative of an ethanol concentration of the fuel. It should be understood that in certain embodiments, the ethanol concentration may be determined directly from the measured capacitance or capacitance-related output signal  117 , without first calculating an actual dielectric constant value. 
         [0025]    The processing circuit  116  includes a current-to-voltage (CTV) converter  122  responsive to the induced current signal  106  for converting the induced current signal  106  into a voltage. The CTV converter  122  includes an operational amplifier  128  and a feedback element  130 , having any one of a number of configurations known in the art. In the illustrated embodiment, the non-inverting input terminal of the op amp  128  is connected to ground, while the inverting input terminal is connected to the node  108  to receive the induced current signal  106 . Additionally, the inverting input terminal and the output terminal of the op amp  128  are electrically connected by way of the feedback element  130 . 
         [0026]    The element  130  is formed with a resistor and a capacitor in parallel. The resistor dominates the feedback impedance. The capacitance with very small value is used to reduce feedback noise but not limited phase lag. 
         [0027]    Through the foregoing described arrangement, the op amp  128  converts the input current signal into an induced voltage signal  132 . The induced voltage signal  132  is a sinusoid voltage with a phase angle identical to the induced sinusoidal current going through the sensing element  60 . It should be understood that at the time when the input excitation voltage signal  102  is zero, the real component of the induced current through the sensing element  60  is also equal to zero. Accordingly, the CTV  122  output signal  132  reflects only the imaginary component of the induced current, now voltage, signal. 
         [0028]    The processing circuit  116  further includes an analog switch circuit, such as a single-port-double-throw (SPDT) analog switch  124 , which will be described immediately below in greater detail. The processing circuit  116  further includes an output DC amplifier  126 , which is configured to amplify the output of the analog switch  124  to produce the output signal  117 . The DC amplifier  126  has a low input bias current. 
         [0029]      FIG. 3  shows analog switch  124  in greater detail. The switch  124  has an input terminal  134 , an output terminal  136  and a sample-and-hold arrangement, such as a capacitor  138  connected to a common node  140 . The switch  124  is controlled by the trigger signal  112  in a manner to be described. In a constructed embodiment, when the trigger signal  112  is HIGH, the switch  124  is in a first state, in which the common node  140  is connected to the input terminal  134  to thereby track the changes in the induced voltage signal  132 . Accordingly, the sample-and-hold capacitor  138  is charged with and thus tracks the CTV output signal  132 . The switch  124  further includes a second state. When the trigger signal  112  goes LOW due to a zero-crossing detection by detector  110 , the trigger signal  112  is operative to control the switch  124  into the second state. In the second state, the common node  140  is switched from the input terminal  134  to the output terminal  136 , and the capacitor  142  is charged with the tracked voltage which is in turn electrically-connected to the DC amplifier  126 . Due to the low input bias current of the DC amplifier  126 , the charge on the capacitor  142  can maintain a relatively constant voltage, which establishes the sampled induced voltage signal  132  on the output terminal  136 . It is preferred that the sample and hold capacitor  138  has larger capacitance value than the capacitor  142 . The voltage on the output terminal  136  is substantially identical to the CTV output signal  132  instantaneously after the zero crossing event. The DC amplifier  126  amplifies this output, which is directly proportional to the imaginary component of the sinusoidal current going through the sensing element  60 . 
         [0030]    It should be appreciated that the sample-and-hold capacitor  138  develops a DC voltage output signal after the zero-crossing event, and thus no rectifiers, filters and the like are necessary, as may have been the case in the prior art where the magnitude signal constituted a sinusoidal signal. This sampling approach provides for a simplified circuit design. 
         [0031]      FIG. 4  is a timing diagram illustrating the operation of the present invention. As can be seen, when the excitation voltage signal  102  crosses zero volts at near time t 1 , the trigger signal  112  transition from a logic HIGH state to a logic LOW state. Between time zero and time t 1 , the output of the sample and hold capacitor  138  at node  140  is coupled to and thus tracks the changes of the CTV output signal  132  (i.e., the two traces overlay each other). However, at time t 1 , the trigger signal  112  controls the switch  124  to disconnect the common node  140  from the CTV output signal  132 . At the point of disconnection, the voltage on the output terminal  136  “holds” its level as it existed at the time the trigger signal  112  transitioned HIGH-to-LOW. As described above, since the time where the excitation voltage signal  102  is at zero volts is where the real (resistive) component of the total complex conductivity (impedance) is also zero, the CTV output signal  132  at that time reflects only the imaginary capacitive component. This signal is transferred to the output terminal  136  and is subsequently amplified by the DC amplifier  126 . 
         [0032]    In sum, the present invention overcomes some of the complexity issues associated with prior approaches. The imaginary component of the total complex conductivity is that which corresponds to the capacitive part of the complex load formed by the fuel and the sensing element. In prior approaches, the imaginary conductivity could be calculated using equation (1) below when total conductivity and the phase angle are known; however, this decomposition was somewhat complex. 
         [0000]        I   m (conductivity)=sin(phase_angle)*total_conductivity   (1) 
         [0033]    where the phase_angle is the calculated or measured phase angle and the total_conductivity is the determined magnitude. 
         [0034]    However, the present invention improves upon prior approaches when only the imaginary part is desired. In particular, the present invention provides for synchronizing the conductivity current sampling time at the point where the excitation voltage equals to zero. At this time, the contribution of the real part (resistive) to the total conductivity is zero, leaving just the imaginary part, which can be measured directly without the need for complex decomposition processing. This results in a greatly simplified system and method for determining a property of a liquid, as explained above. 
         [0035]      FIG. 5  shows the known circuit design for a liquid property sensor, as described in the Background, and as disclosed in U.S. Pat. No. 6,693,444 issued to Lin et al. entitled CIRCUIT DESIGN FOR LIQUID PROPERTY SENSOR. This circuit develops both a magnitude signal indicative of total conductivity as well as a phase signal indicative of the phase angle between the excitation signal and the current through the sensing element. For completeness, a description of this circuit will be given below. 
         [0036]    As described in the Background, it is know to provide a complex impedance circuit that is based on a single excitation frequency. This apparatus provides both a magnitude and phase angle. In this regard,  FIG. 5  is a block diagram of a sensor apparatus  150  that can perform this impedance determination. The sensing element  160  of the sensor  150  comprises two spaced electrodes, an excitation plate  160   a  and a sensing plate  160   b , both made of a conductive material. The sensing element  160  is submerged in the fuel and excited by a sinusoidal wave generator  152 . The sinusoidal wave generator  152  generates a sinusoidal voltage centered at the voltage Vdd/2. By example, the peak-to-peak amplitude is around 4 volts. The sinusoidal voltage is at a single frequency in the range of 10 kHz to 100 kHz. If the generator is a single stage sine wave generator  152 , the voltage is first filtered through a standard low pass filter  154  to filter out high order harmonics. Alternatively, of course, a dual stage sine wave generator  152  can be used and the low pass filter  154  omitted. 
         [0037]    The filtered voltage feeds through a voltage divider tied to Vdd/2. The resulting voltage signal provides a temperature reference voltage  155  to a switch  166  The temperature reference voltage  155 , and its use with the switch  166 , is discussed further herein. The filtered voltage also provides an excitation signal to the sensing element  160  through the shielded cable  122  at node  158 . Specifically, the filtered voltage flows through a DC block capacitor  156 , and the resulting excitation signal reaches the excitation plate, or electrode,  160   a  of the sensing element  160 . Node  158  brings the DC voltage of the excitation plate  160   a  of the sensing element  160  down to ground through a grounding resistor. 
         [0038]    The control circuit  148  receives the excitation signal from node  158  and supplies it as a reference input excitation signal  175  for a pulse width modulated (PWM) generator  176 , discussed herein. 
         [0039]    The control circuit  148  receives the current induced on the sensing element  160  from the sensing electrode  160   b  through the shielded cable  122 . Preferably, the sensing plate, or electrode,  160   b  of the sensing element  160  is grounded through a resistor to bring the DC components of this induced signal to ground. Together with the ground provided for the excitation plate  160   a  at node  158 , this ground assures that the signals supplied to the remainder of the control circuit  148  have no DC components. Also, and as shown in  FIG. 5 , the shield or the shielded cable  122  is preferably brought to ground, optionally through a resistor (not shown). As additional protection against DC components in the induced signal, a series-connected DC blocking capacitor  162  filters the induced signal prior to it being supplied to the inverting input of an operational amplifier (op amp)  164   a  configured as a current-to-voltage converter  164 . 
         [0040]    In the current-to-voltage converter  164 , the inverting input of the op amp  164   a  is raised to Vdd/2 through a resistor, as is the non-inverting input of the op amp  164   a . Feedback is supplied through a feedback impedance  164   b , wherein either the reactive component or the resistive component of the feedback impedance  164   b  is minimized. Preferably, the feedback impedance  164   b  provides the op amp  164   a  with a variable gain such that the resolution of the output signal MAGNITUDE is adjustable by changing the feedback impedance. Ideally, the output of the converter  164  is a sinusoidal voltage centered at, for example, 2.5 volts. Depending upon the characteristics of the fuel, however, the op amp  164   a  can saturate, and the resolution of the signal MAGNITUDE, discussed herein, diminishes. One characteristic affecting the resolution of the signal is the ethanol content. 
         [0041]    In the preferred embodiment, the feedback impedance  164   b  comprises a plurality of parallel complex impedances enabled by a gain control signal GAIN. By example, four complex impedances are connected to four outputs of a digital switch, and each complex impedance includes a large resistance value in parallel with a small capacitance value. The gain control signal GAIN is a digital signal generated by the engine controller  118  or a microcontroller (not shown) of the control electronics  148 , here [0:0] to [1:1]. Whichever controller receives the output MAGNITUDE sends the signal GAIN to the digital switch, adjusting the gain of the op amp  164   a  until the output MAGNITUDE reaches the desired resolution. Where the fuel has a large capacitance, a small gain is desirable; where the fuel has a small capacitance, a large gain is desirable. 
         [0042]    The output of the current-to-voltage converter  164  is a sinusoidal voltage signal  165  centered at, for example, 2.5 volts, and representative of the complex impedance of the fuel. The sinusoidal voltage signal  165 , like the temperature reference voltage  155 , is preferably fed into the switch  166 . The switch  166  can be an analog switch, such as ADG419 from Analog Devices, Inc. of Norwood, Mass., which receives a sampling signal SELECT from the engine controller  118  or a microcontroller (not shown) of the control electronics  148 . The sampling signal SELECT determines which of the sinusoidal voltage signal  165  and the temperature reference voltage  155  are used to calculate the output signal MAGNITUDE. This provides a means of correcting the output signal MAGNITUDE for temperature variations of the circuit board on which the control electronics  148  are mounted. 
         [0043]    More specifically, the temperatures to which the sensor  150  is exposed vary significantly with operation of the vehicle in which the sensor  150  is installed. Circuit board temperatures can range, for example, from −40° C. to 125° C. Normally, the sampling signal SELECT is such that the sinusoidal voltage signal  165  passes through and is used to determine the output signal MAGNITUDE. The output signal MAGNITUDE is a DC voltage used by the controller in a lookup table, for example, to determine the impedance magnitude of the complex impedance. Testing shows, however, that signal drops for a nominal magnitude of 2 volts can be 10% or more as the temperature increases. The present invention addresses this problem by, at specific predetermined intervals, sending a sampling signal SELECT that enables the switch  166  to pass the temperature reference voltage  155  on to the remainder of the control electronics  148  that determines the output signal MAGNITUDE. This output signal MAGNITUDE is compared to the expected magnitude based upon the value of the voltage reference Vdd/2. A ratio, or adjustment factor, of the output signal MAGNITUDE developed from the temperature reference voltage  155  to the expected voltage is used to adjust the output signal MAGNITUDE based upon the sensed sinusoidal voltage signal  165 . In this manner, the output signal MAGNITUDE is adjusted for temperature variation prior to using it to determine the impedance magnitude of the complex impedance. 
         [0044]      FIG. 5  shows one circuit design that can detect the peak of the sinusoidal voltage output of the switch  166 , whether it is the sensed sinusoidal voltage signal  165  or the temperature reference voltage  155 . First, the signal is rectified by a standard full wave rectifier  168 . After passing through a buffer  170 , the signal is filtered through a low pass filter  172  to remove its AC components. The resulting DC signal is then fed through a differential amplifier  174 , which sends the amplified DC signal, output signal MAGNITUDE, to a microcontroller, such as the engine controller  118 . The engine controller  118  then adjusts the output signal MAGNITUDE by the last calculated adjustment factor if the output signal MAGNITUDE is based upon the sensed sinusoidal voltage signal  165 , or a new adjustment factor is determined if the output signal MAGNITUDE is based upon the temperature reference voltage  155 . 
         [0045]    Optionally, the actual magnitude of the complex impedance can be determined from this voltage output signal MAGNITUDE. To do this, the engine controller  118  compares the output signal MAGNITUDE to values on a look up table determined in prior calibration experiments wherein the look up table correlates voltage outputs to impedance magnitudes. Alternately, a mathematical relationship between these two variables can be developed and used by the engine controller  118  to determine the impedance magnitude from the output signal MAGNITUDE. 
         [0046]    The output of the current-to-voltage converter  164 , which is representative of the complex impedance of the fuel, takes two paths. As described above, the sinusoidal voltage signal  165  is supplied to a peak detector, or any kind of an AC amplitude to DC converter that detects the magnitude of the peak of the signal. Second, the sinusoidal voltage signal  165  is supplied to the PWM generator  176 , which compares that voltage signal  165  to the reference input excitation signal  175  to determine the phase of the complex impedance. A multitude of circuits can determine this phase from the two input signals; one is shown in  FIG. 5 . 
         [0047]    The PWM generator  176  of  FIG. 5  includes two comparators  176   a  and  176   b  and a pulse-width modulator circuit  176   c . In the example, the sinusoidal voltage signal  165  is a sinusoidal voltage centered at 2.5 volts. It is supplied to the non-inverting input of the comparator  176   a , while the inverting input of the comparator  176   a  is at Vdd/2. The output of the comparator  176   a  is a square wave  177  from 0 to 5 volts with a frequency corresponding to that of the sinusoidal voltage signal  165 . The reference input excitation signal  175  is a sinusoidal voltage centered at 0 volts at the same frequency as the sinusoidal voltage signal  165 . However, the sinusoidal voltage signal  165  is offset in phase from the reference input excitation signal  175 , where the offset corresponds to the phase of the impedance between the node  158  and the output of the op amp  164   a  of the current-to-voltage converter  164 . The reference input excitation signal  175 , like the sinusoidal voltage signal  165 , is similarly supplied to the non-inverting input of a comparator  176   b , while the inverting input of the comparator  176   b  is at ground. The output of the comparator  176   b  is a square wave  179  from 0 to 5 volts with a frequency corresponding to that the reference input excitation signal  175  and with the same phase offset from the sinusoidal voltage signal  165 . The two square waves  177  and  179  are provided to two field-effect transistors (FET) of a pulse-width modulator circuit  176  comprising three FETs. More specifically, each of the two square waves  177 ,  179  is provided as an input to the gate of a corresponding FET  176   d ,  176   e . The source of each of the three FETs  176   d - f  is grounded, while the drain of each of the three FETs  176   d - f  is raised to Vdd through a resistive load. The output voltage at the drain of the FET  176   d  receiving the square wave  177  is the input voltage signal for the gate of the third FET  176   f , while the output voltage of the drain of the FET  176   e  receiving the square wave  179  is tied to the output voltage of the drain of the third FET  176   f . Thus, the output of the pulse-width modulator circuit  176   c , and of the PWM generator  176 , is a square wave from 0 to 5 volts with a duty cycle based upon the difference in phase, or the phase offset, of the square wave  177 , representing the induced signal, and the square wave  179 , representing the excitation signal. 
         [0048]    The output of the PWM generator  176  is passed through a conventional low pass filter with a fixed gain  178 . The resulting output signal PHASE is a square wave with a duty cycle ranging from 0%-50%, which is provided to the same controller as the output signal MAGNITUDE, such as the engine controller  118 . The controller  118  calculates the duty cycle according to conventional methods. Through prior calibration, another look up table can be provided in the engine controller  118  whereby a duty cycle of 0%-50% corresponds to a phase of the complex impedance of 0°-180°. Once the controller  118  has the duty cycle of the output signal PHASE, it can use the look up table to determine the phase of the complex impedance. Of course, as with the calculation of the magnitude of the complex impedance, a mathematical relationship governing the relationship of the output signal PHASE to the phase of the complex impedance can be developed from the prior calibration experiments and used instead of the look up table. Given the complex output comprising the magnitude and the phase outputs, the microcontroller or engine controller  118  can determine the resistance and capacitance of the fuel by a simple calculation. 
         [0049]    Thus is presented in  FIG. 5  a sensor design can measure capacitance down to the picofarad range and measure magnitude and phase difference using a single excitation frequency in the range of 10-100 kHz. A simple calculation gives the precise measurements of resistance and capacitance. 
         [0050]    While the invention has been described in connection with what is presently considered to be the most practical and preferred embodiment, it is to be understood that the invention is not to be limited to the disclosed embodiments but, on the contrary, is intended to cover various modifications and equivalent arrangements included within the spirit and scope of the appended claims, which scope is to be accorded the broadest interpretation so as to encompass all such modifications and equivalent structures as is permitted under the law.