Abstract:
Provided is a wideband receiver that has a smaller area and consumes less power and can prevent harmonic mixing occurring due to an increase in the number of communications systems using wideband. A wideband receiver according to an aspect of the invention may include: an front-end unit receiving and performing low-pass filtering on a wideband input signal in a continuous-time domain; and a down-conversion unit sampling and holding an output signal of the front-end unit according to a local oscillator signal and performing low-pass filtering on the output signal in a discrete tie domain.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application claims the priority of Korean Patent Application Nos. 10-2009-0127546 filed on Dec. 18, 2009, and 10-2010-0035945 filed on Apr. 19, 2010, in the Korean Intellectual Property Office, the disclosures of which are incorporated herein by reference. 
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to wideband receivers, and more particularly, to a wideband receiver that has a smaller area and consumes less power and can prevent harmonic mixing occurring due to an increase in the number of communications systems using wideband. 
     2. Description of the Related Art 
     An application using wideband, such as a digital TV, frequently undergoes harmonic mixing as shown in  FIGS. 1A ,  1 B and  1 C. 
     As shown in  FIG. 1A , as for wideband applications, undesirable signals are present at frequencies three or five times the magnitude of a carrier frequency of an input signal (an RF signal). By a frequency three or five times the magnitude of a frequency of a local oscillator (LO) signal, shown in  FIG. 1B , these signals are moved down to baseband frequencies together with a desired signal when the frequency of the desired signal is down-converted to baseband by a down-conversion mixer, which is illustrated in  FIG. 1C . 
     In order to solve the above-described problem of harmonic mixing, in the related art, a dual conversion receiver has been used. 
     As shown in  FIG. 2 , a dual conversion receiver amplifies an input signal RFin by a wideband low-noise amplifier (hereinafter, referred to as an “LNA”)  211  and up-converts the amplified input signal to high intermediate frequency (IF) by an up-conversion mixer  212 . The input signal being up-converted is filtered through a narrowband surface acoustic wave (SAW) filter  213 , is then moved to low IF by down-conversion mixers  221  and  222 , and is finally output via IF variable gain amplifiers (hereinafter, referred to as “VGAs”)  223  and  224  and low pass filters (hereinafter, referred to as “LPFs”)  225  and  226 . 
     However, the dual conversion receiver having the above-described configuration and performing the above-described operation has various problems as follows. 
     First, the use of additional external components, such as the narrowband SAW filter  213 , causes an increase in manufacturing costs. 
     Besides, unlike existing receivers, the dual conversion receiver requires two frequency synthesizers  214  and  228  and two or more mixers, that is, the up-conversion mixer  212  and the down-conversion mixers  221 , and  222 , thereby increasing the size of the receiver and the amount of power being consumed. 
     In order to solve these problems of the dual conversion receiver, in the related art, another receiver that includes a harmonic suppression mixer having a plurality of mixers connected in parallel with each other was additionally proposed. 
     However, since the harmonic suppression mixer has a larger area and consumes more power than existing mixers, the area and power consumption of the receiver having the harmonic suppression mixer therein are therefore increased. Besides, the harmonic suppression mixer requires a plurality of multi-phase local oscillator signals, and accurate phase differences need to exist between the plurality of local oscillator signals. For this reason, additional circuits need to be added in order to control the phase differences between the plurality of local oscillator signals, which result in the receiver having a higher area and high power consumption. 
     SUMMARY OF THE INVENTION 
     An aspect of the present invention provides a wideband receiver having new architecture that has a smaller area and consumes less power and can prevent harmonic mixing occurring due to an increase in the number of communications systems using wideband. 
     According to an aspect of the present invention, there is provided a wideband receiver including: an front-end unit receiving and performing low-pass filtering on a wideband input signal in a continuous-time domain; and a down-conversion unit sampling and holding an output signal of the front-end unit according to a local oscillator signal and performing low-pass filtering on the output signal in a discrete tie domain. 
     The front-end unit may include: a wideband low-noise amplifier receiving and amplifying the wideband input signal; and at least one tunable low pass filter changing cutoff frequency according to a frequency of the wideband input signal and performing low-pass filtering on the wideband input signal. 
     The at least one tunable low pass filter may include: first and second resistors connected in series between an input terminal and an output terminal; a first capacitor connected between a ground and a contact point between the output terminal and the second resistor; a second capacitor and an output resistor connected between the ground and a contact point between the first resistor and the second resistor; and a buffer connected between the output terminal and a contact point between the second capacitor and the output resistor. 
     The buffer may be configured as an operational amplifier having a gain of 1. 
     The at least one tunable low pass filter may change the cutoff frequency by changing a device value of at least one of the first and second resistors and the first and second capacitors. 
     The at least tunable low pass filter further may include: a first transistor of a first conductivity type having a gate connected to the input terminal and a source to which a driving voltage is applied; a first transistor of a second conductivity type having a gate and a drain connected in common to a drain of the first transistor of the first conductivity type; and a second transistor of a first conductivity having a gate and a drain connected in common to the output terminal and a source connected to the driving voltage terminal. 
     The buffer may be configured as a second transistor of a second conductivity type having a gate connected to the first capacitor, a drain connected to the drain of the second transistor of the first conductivity type, and a source connected to the second capacitor. 
     The down-conversion unit may include: a clock generator generating a clock; a phase shifter shifting a clock with a phase difference of 90° to thereby generate the local oscillator signal required to restore an I/Q signal; two sample and hold circuits sampling and holding the output signal of the front-end unit according to the local oscillator signal in the discrete-time domain, down-converting the output signal of the front-end unit to baseband, and converting the output signal into a signal in the discrete-time domain; and two discrete-time low pass filters performing low-pass filtering on respective outputs of the two sample and hold circuits in the discrete-time domain. 
     Each of the two sample and hold circuits may include: a first transistor having a drain connected to a first input terminal and a gate to which a local oscillator signal is input; a second transistor having a drain connected to a second input terminal and a gate to which the local oscillator signal is input; a third transistor having a drain connected to a source of the first transistor, a gate to which an inverted local oscillator signal is input, and a source connected to an output terminal; a fourth transistor having a drain connected to a source of the second transistor, a gate to which the inverted local oscillator signal is input, and a source connected to a bias voltage; a fifth transistor having a drain connected to the output terminal and a gate to which the local oscillation signal is input; a first capacitor connected between the source of the first transistor and the source of the second transistor; a second capacitor connected between the output terminal and a source of the fifth transistor; and a third capacitor connected between the second capacitor and a ground. 
     The two sample and hold circuits may receive respective output signals of the front-end unit in the form of a differential signal pair through the first and second input terminals. 
     The clock generator may vary a frequency of the clock. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The above and other aspects, features and other advantages of the present invention will be more clearly understood from the following detailed description taken in conjunction with the accompanying drawings, in which: 
         FIGS. 1A ,  1 B, and  1 C are views illustrating harmonic mixing; 
         FIG. 2  is a view illustrating a dual conversion receiver according to the related art; 
         FIG. 3  is a view illustrating a wideband receiver according to an exemplary embodiment of the present invention; 
         FIG. 4  is a view illustrating a tunable LPF according to an exemplary embodiment of the present invention; 
         FIG. 5  is a view illustrating a tunable LPF according to another exemplary embodiment of the present invention; 
         FIG. 6  is a view illustrating a tunable LPF according to another exemplary embodiment of the present invention; 
         FIGS. 7A and 7B  are graphs illustrating the filter characteristics of the tunable LPF of  FIG. 5 ; 
         FIGS. 8A and 8B  are graphs illustrating the filter characteristics of the tunable LPF of  FIG. 6 ; and 
         FIG. 9  is a view illustrating a sample and hold circuit according to an exemplary embodiment of the present invention. 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     As the present invention allows for various changes and numerous embodiments, particular embodiments will be illustrated in drawings and described in detail in the written description. 
     However, this is not intended to limit the present invention to particular modes of practice, and it is to be appreciated that all changes, equivalents, and substitutes that do not depart from the spirit and technical scope of the present invention are encompassed in the present invention. 
     While such terms as “first” and “second,” etc., may be used to describe various components, such components must not be limited to the above terms. The above terms are used only to distinguish one component from another. For example, a first component may be referred to as a second component without departing from the scope of the rights of the present invention, and likewise, a second component may be referred to as a first component. 
     When a component is mentioned to be “connected” to or “accessing” another component, this may mean that it is directly connected to or accessing the other component, but it is to be understood that another component may exist in-between. On the other hand, when a component is mentioned as being “directly connected” to or “directly accessing” another component, it is to be understood that there are no other components in-between. 
     The terms used in the present application are merely used to describe particular embodiments, and are not intended to limit the present invention. An expression used in the singular encompasses the expression of the plural, unless it has a clearly different meaning in the context. In the present application, it is to be understood that the terms such as “including” or “having,” etc., are intended to indicate the existence of the features, numbers, operations, actions, components, parts, or combinations thereof disclosed in the specification, and are not intended to preclude the possibility that one or more other features, numbers, operations, actions, components, parts, or combinations thereof may exist or may be added. 
     Unless otherwise defined, all terms used herein, including technical or scientific terms, have the same meanings as those generally understood by those with ordinary knowledge in the field of art to which the present invention belongs. Such terms as those defined in a generally used dictionary are to be interpreted to have meanings equal to the contextual meanings in the relevant field of art, and are not to be interpreted to have ideal or excessively formal meanings unless clearly defined in the present application. 
     Embodiments of the present invention will be described below in detail with reference to the accompanying drawings, where those components are rendered the same reference number that are the same or correspond to, regardless of the figure number, and redundant explanations are omitted. 
       FIG. 3  is a view illustrating a wideband receiver according to an exemplary embodiment of the invention. 
     Referring to  FIG. 3 , a wideband receiver according to this embodiment includes an front-end unit  310  and a down-conversion unit  320 . The front-end unit  310  receives and performs low-pass filtering on a wideband input signal in the continuous-time domain. The down-conversion unit  320  samples and holds an output signal from the front-end unit  310  according to a local oscillator signal and performs low-pass filtering on the output signal in the discrete-time domain. 
     The front-end unit  310  includes a wideband LNA  311  and a tunable low pass filter (hereinafter, referred to as a “tunable LPF”)  312 . The wideband LNA  311  receives and amplifies the wideband input signal RFin. The tunable LPF  312  can change cutoff frequency and performs low-pass filtering on the input signal according to the cutoff frequency in the continuous-time domain. 
     Here, the tunable LPF  312  may be a high pass filter in order to remove signals present at frequencies higher than a desired frequency to thereby prevent harmonic mixing. Furthermore, since the amplitude of an undesirable signal cannot be reduced to a desired level or less unless the cutoff frequency (or 3 dB frequency) is changed according to the frequency of the input signal RFin, the cutoff frequency can be changed according to the frequency of the input signal RFin. 
     The down-conversion unit  320  includes a clock generator  321 , a phase shifter  322 , two sample and hold circuits (hereinafter, referred to as “SAH circuits”)  323  and  324 , and discrete-time low pass filters (hereinafter, referred to as “DT LPFs”)  325  and  326 . The clock generator  321  generates a clock. The phase shifter  322  shifts the clock with a phase difference of 90° to thereby generate a local oscillator signal LO required to restore an I/Q signal. The SAH circuits  323  and  324  each sample and hold a signal, being output from the front-end unit  310 , according to the local oscillator signal LO in the discrete-time domain, down-convert the signal to baseband, and then convert the signal to a signal in the discrete-time domain. The DT LPFs  325  and  326  perform low-pass filtering on respective outputs from the SAH circuits  323  and  324  in the discrete-time domain. Here, according to the known art, the DT LPFs  325  and  326  may be IIR (infinite impulse response) filters or FIR (finite impulse response) filters. A detailed description thereof will be omitted. 
     That is, as the down-conversion unit  320  uses the SAH circuits  323  and  324  and the DT LPFs  325  and  326  that are operable in the discrete-time domain, the down-conversion unit  320  can be operated in the discrete-time domain. Here, the operating characteristics of the down-conversion unit  320  can be varied by changing the frequency of the clock, being generated by the clock generator  321 . In particular, the filtering characteristics of the DT LPFs  325  and  326  can be easily changed according to the frequency of the clock, so that the wideband receiver according to the exemplary embodiment of the invention can be used in various manners in another applications as well as digital TVs. 
     Furthermore, the DT LPFs  325  and  326  also serve as decimation filters used to reduce respective sampling frequencies of the SAH circuits  323  and  324 . In comparison with LPFs operating in the continuous-time domain, the DT LPFs  325  and  326  are less sensitive to process, voltage and temperature variations, thereby increasing the reliability of the operation of the wideband receiver. 
       FIG. 4  is a view illustrating a tunable LPF according to an exemplary embodiment of the invention. 
     As shown in  FIG. 4 , a tunable LPF  312 - 1  has a structure of a Sallen-Key filter. More specifically, the tunable LPF  312 - 1  includes first and second resistors R 1  and R 2  connected in series between an input terminal IN and an output terminal OUT, a first capacitor C 1  connected between a ground and a contact point between the output terminal OUT and the second resistor R 2 , a second capacitor C 2  and an output resistor Rout connected in series between the ground and a contact point between the first resistor R 1  and the second resistor R 2 , and a buffer B connected between the output terminal OUT and a contact point between the second capacitor C 2  and the output resistor Rout. Here, the buffer B may be configured as an operational amplifier having a gain of “1”. 
     The tunable LPF having the above-described configuration determines the cutoff frequency fcutoff according to Equation 1. 
     
       
         
           
             
               
                 
                   
                     f 
                     cutoff 
                   
                   = 
                   
                     1 
                     
                       2 
                       ⁢ 
                       π 
                       ⁢ 
                       
                         
                           C 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           1 
                           ⁢ 
                           C 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           2 
                           ⁢ 
                           R 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           1 
                           ⁢ 
                           R 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           2 
                         
                       
                     
                   
                 
               
               
                 
                   Equation 
                   ⁢ 
                   
                       
                   
                   ⁢ 
                   1 
                 
               
             
           
         
       
     
     Referring to Equation 1, it can be seen that the cutoff frequency fcutoff of the tunable LPF is determined by device values of the first and second resistors R 1  and R 2  and the first and second capacitors C 1  and C 2 . 
     In the present invention, therefore, at least one of the first and second resistors R 1  and R 2  and the first and second capacitors C 1  and C 2  is realized as an array or variable device, and a device value thereof is changed according to the frequency of the input signal RFin, so that the cutoff frequency fcutoff is finally changed. 
     Generally, when both the input and output of the LPF are voltages, linearity in a low frequency band may be reduced. In particular, since the V-I characteristic of the transistor is not linear, if V-I conversion continues to be performed, the linearity in the low frequency band can be further reduced. 
     Thus, in the present invention, as shown in  FIG. 5 , components are added in order to convert voltages at the input terminal IN and the output terminal OUT of the tunable LPF, shown in  FIG. 4 , into currents, so that the input of the LPF is changed into a current. 
       FIG. 5  is a view illustrating a tunable LPF according to another exemplary embodiment of the invention.
         Referring to  FIG. 5 , a tunable LPF  312 - 2  includes the first through third output resistors R 1 , R 2 , and Rout, the first and second capacitors C 1  and C 2 , and the buffer B as shown in  FIG. 4 . The tunable LPF  312 - 2  further includes a first PMOS transistor PM 1 , a first NMOS transistor NM 1 , an output resistor Rout, and a second PMOS transistor PM 2 . The first PMOS transistor PM 1  has a gate connected to an input terminal IN and a source to which a driving voltage Vdd is applied. The first NMOS transistor NM 1  has a gate and a drain connected in common to a drain of the first PMOS transistor PM 1 . The output resistor Rout is connected to a source of the first NMOS transistor NM 1 . The second PMOS transistor PM 2  has a gate and a drain connected in common to an output terminal OUT and a source connected to a driving voltage Vdd terminal.       

     That is, as the first PMOS transistor PM 1  and the first NMOS transistor NM 1  are added to the input terminal IN of the tunable LPF  312 - 2 , shown in  FIG. 5 , and the second PMOS transistor PM 2  is added to the output terminal OUT thereof, the tunable LPF  312 - 2  converts an input voltage and an output voltage into an input current and an output current, respectively, by using the added components. 
     Furthermore, as shown in  FIG. 5 , the buffer B, shown in  FIG. 4 , may be realized as a second NMOS transistor NM 2  having a gate connected to the first capacitor C 1 , a drain connected to the drain of the second PMOS transistor PM 2 , and a source connected to the second capacitor C 2 . Basically, a transistor can be driven using power smaller than that of an operational amplifier, in the case that a buffer of the operational amplifier is replaced with a transistor, the amount of power being consumed by the tunable LPF can be reduced. 
     Furthermore, when a rejection ratio of the tunable LPF having the configuration as shown in  FIGS. 4 and 5  is not high enough, a plurality of tunable LPFs are connected in series with each other as shown in  FIG. 6 , so that a rejection ratio with respect to an undesirable signal can be improved. 
       FIG. 6  is a view illustrating a tunable LPF according to another exemplary embodiment of the invention. 
     Referring to  FIG. 6 , a tunable LPF  312 - 3  has a plurality of tunable LPFs  312 - 2  as shown in  FIG. 4  or  FIG. 5 , connected in series with each other. 
     When the tunable LPFs, shown in  FIG. 6 , are configured using the tunable LPFs  312 - 2  as shown in  FIG. 5 , a second PMOS transistor PM 2  of the tunable LPF  312 - 2 , provided at a front stage, and a first PMOS transistor PM 1  of the tunable LPF  312 - 2 , provided at a rear stage, are connected in a current mirror configuration. As a result, an output current of the tunable LPF  312 - 2 , provided at the front stage, is thereby applied as an input current of the tunable LPF  312 - 2 , provided at the rear stage. 
       FIGS. 7A and 7B  are views illustrating the filter characteristics of the tunable LPF as shown in  FIG. 5 .  FIGS. 8A and 8B  are views illustrating the filter characteristics of the tunable LPF as shown in  FIG. 6 . 
     As shown in  FIGS. 7A and 7B  and  8 A and  8 B, a tunable LPF according to an exemplary embodiment of the invention can optionally control the cutoff frequency by changing device values of resistors and capacitors. That is, as shown in  FIGS. 7A and 7B  and  8 A, a cutoff frequency may be set to 80 MHz. Alternatively, as shown in  FIGS. 7A and 7B  and  8 B, a cutoff frequency may be set to 1 GHz. 
     Through a comparison between the drawings of  FIGS. 7A and 7B  and  FIGS. 8A and 8B , it can be seen that the tunable LPFs, shown in  FIG. 6 , have a higher rejection ratio than the tunable LPF, shown in  FIG. 5 . That is, by connecting the tunable LPFs, as shown in  FIG. 5 , in series with each other, a rejection ratio with respect to an undesirable signal can be increased. 
       FIG. 9  is a view illustrating an SAH circuit according to an exemplary embodiment of the invention. 
     As shown in  FIG. 9 , each of the SAH circuits  323  and  324  includes a first transistor M 1 , a second transistor M 2 , a third transistor M 3 , a fourth transistor M 4 , a fifth capacitor M 5 , a first capacitor C 1 , a second capacitor C 2 , and a third capacitor C 3 . The first transistor M 1  has a drain connected to a first input terminal INP and a gate to which a local oscillator signal LO+ is input. The second transistor M 2  has a drain connected to a second input terminal INN and a gate to which a local oscillator signal LO+ is input. The third transistor M 3  has a drain connected to a source of the first transistor M 1 , a gate to which an inverted local oscillator signal LO− is input, and a source connected to an output terminal OUT. The fourth transistor M 4  has a drain connected to a source of the second transistor M 2 , a gate to which the inverted local oscillator signal LO− is input, and a source connected to a bias voltage VBIAS terminal. The fifth transistor M 5  has a drain connected to the output terminal OUT and a gate to which the local oscillator signal LO+ is input. The first capacitor C 1  is connected between the source of the first transistor M 1  and the source of the second transistor M 2 . The second capacitor C 2  is connected between the output terminal OUT and a source of the fifth capacitor M 5 . The third capacitor C 3  is connected between the second capacitor C 2  and a ground. Here, the SAH circuit has a differential structure receiving an output of the tunable LPF  312  and a local oscillator signal from the phase shifter  322  in the form of a differential signal pair. 
     Hereinafter, the operation of the SAH circuit will be described. 
     First, when a local oscillator signal pair consisting of a local oscillator signal LO+ and an inverted local oscillator signal LO− and having a first value is applied (for example, a local oscillator signal LO+ has a high level and an inverted local oscillator signal LO− is a low level), the first, second, and fifth capacitors M 1 , M 2 , and M 5  are turned on, and the third and fourth transistors M 3  and M 4  are turned off. Both ends of the first capacitor C 1  are then connected to the first and second input terminals INP and INN, respectively, a signal value of the input signal pair is stored in the first capacitor C 1 . 
     Subsequently, when a local oscillator signal pair consisting of a local oscillator signal LO+ and an inverted local oscillator signal LO− and having a second value is applied (for example, the local oscillator signal LO has a low level, and the inverted local oscillator signal LO− has a high level), the first, second, and fifth capacitors M 1 , M 2 , and M 5  are turned off, and the third and fourth transistors M 3  and M 4  are turned on. The signal value of the input signal pair, stored in the first capacitor C 1 , is finally output to the third transistor M 3  and the second capacitor C 2 . 
     That is, the SAH circuit, as shown in  FIGS. 8A and 8B , samples the signal value of the input signal pair in a half period of the local oscillator signal pair consisting of the local oscillator signal LO+ and the inverted local oscillator signal LO−, and outputs the sampled signal value to the output terminal OUT. 
     As set forth above, according to exemplary embodiments of the invention, a wideband receiver receives and performs low-pass filtering on a wideband input signal in the continuous-time domain, down-converts the wideband input signal, and performs low-pass filtering on the wideband input signal in the discrete-time domain. Therefore, a large number of PLLs and mixers are not required, so that the wideband receiver has a small area and consumes less power and can prevent harmonic mixing. Furthermore, since a down-conversion unit is operated in the discrete-time domain, the filtering characteristics of the down-conversion unit can be varied by changing the frequency of a clock required to operate the down-conversion unit. 
     While the present invention has been shown and described in connection with the exemplary embodiments, it will be apparent to those skilled in the art that modifications and variations can be made without departing from the spirit and scope of the invention as defined by the appended claims.