Abstract:
A single-wire time-domain reflectometer (TDR) combines the best performance features of prior art “electronic dipsticks” in a high accuracy implementation that allows tank penetration though a small opening. A wire-horn structure is employed to launch TDR pulses onto a single wire transmission line, wherein the horn wires can be flexed inwards so the dipstick structure can be inserted through a small tank opening. Once inside the tank, the horn wires flex to their normal state to provide a controlled reference reflection while simultaneously providing high coupling efficiency to the dipstick. The TDR system determines the fill-level of a tank by measuring the time difference between a reflection created at the wire-horn, which all is at the top of a tank, and a reflection from a material in the tank. The TDR employs automatic time-of-peak (TOP) detectors and incorporates a 2-diode sampler, a low-aberration pulse generator, and a 0.001% accurate timebase.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to pulsed electromagnetic sensors, and more particularly to fluid and material level sensors using time-domain reflectometry (TDR). These sensors can be used for determining or controlling the fill-level of a tank, vat, irrigation ditch, silo, pile,. or conveyor. Also, the present invention can be used as a linear displacement transducer for use in machine control. 
     2. Description of Related Art 
     TDR techniques have been used in the past for measuring the fill-level in a tank. For example, U.S. Pat. No. 3,703,829, Liquid Quantity Gaging System, to Dougherty discloses a time-domain reflectometer (TDR) connected to a coaxial cable, or probe, immersed in a liquid, wherein the time delay of the reflected pulse is a measure of the liquid level in the coaxial probe. The key advantages to coaxial TDR probes are (1) strong reflection amplitudes, which are of particular advantage with low dielectric constant materials, and (2) stilling action, wherein sloshing is less pronounced inside the coaxial probe so steadier measurements can be obtained. On the negative side, coaxial probes are (1) mechanically difficult to fabricate with adequate precision, particularly concerning the centering and support of the open-air center conductor, (2) difficult to cut in custom lengths in the field, (3) difficult to ship in long sections, (4) difficult to join in short segments, (5) susceptible to blockage, and (6) difficult to make flexible for coiling during shipping. 
     A single wire transmission line, or Goubau line, overcomes most of the limitations to the coaxial probe and has been disclosed in U.S. Pat. No. 3,995,212, Apparatus and Method for Sensing a Liquid with a Single Wire Transmission Line, to Ross and U.S. Pat. No. 5,609,059, Electronic Multi-purpose Material Level Sensor, to McEwan. The key advantages to a single wire TDR probe for material level sensing are (1) extreme simplicity, (2) ability to coil the line for shipping (when made of wire), (3) simple custom cutting to length in the field, (4) nearly complete freedom from clogging (material can cling to the line, but generally has little effect), and (5) low cost. 
     A single wire probe requires a means to launch a TDR pulse onto the wire. A horn launcher, as described by Ross, exhibits high launching efficiency and provides a smooth impedance transition between the TDR unit and the high impedance of the single wire transmission line. However, the horn has notable disadvantages: (1) there is an impedance discontinuity that extends along the length of the horn that casts a distributed reflection and creates a potential measurement error, (2) there is no definite reflection to provide a “top-of-tank” reference marker, (3) the horn ends too abruptly at its rim which creates a spurious reflection in the measurement range, (4) the horn is physically large and expensive, and (5) a large opening is needed to insert the horn through, often requiring a large, and therefore expensive, ANSI-rated tank cover. 
     (ANSI is the American National Standards Institute.) 
     A flat plate-type launcher, as described by McEwan in U.S. Pat. No. 5,609,059, creates a strong reflection to indicate the top of the tank, is mechanically simple, and does not require a large tank opening. Its primary disadvantages are (1) the launch point reflection is often too strong, creating pulse aberrations that extend into the measurement range, (2) it has a low launch efficiency relative to the horn, which results in excessively low signal returns from low dielectric constant materials, (3) due to its low launch efficiency, a hot ground condition exists that can propagate pulses backwards onto the outside of the TDR feed cable, creating spurious reflections and ringing. 
     A launcher is needed that combines the best performance features of both the horn and the plate with none of the drawbacks: good coupling efficiency, a controlled-amplitude marker reflection, absence of hot grounds, insertable through a small opening, and low cost. 
     Regardless of whether a coaxial or single wire line is used, it is most desirable to process the reflected pulses with automatic pulse detection techniques that render the measurement independent of pulse amplitude. McEwan, in U.S. Pat No. 5,610,611, High Accuracy Material Level Sensor, discloses a constant fraction discriminator, or CFD, that incorporates a peak detector to automatically set the trigger point on its pulse detectors. While this method eliminates pulse amplitude dependence, it suffers from dynamic errors that can arise in sloshing tanks. The dynamic errors arise from the inability of the peak detector to track rapid decreases in repetitive pulse amplitude. A new automatic pulse detector is needed, and preferably one which also rejects errors caused by low-frequency aberrations in the return signal. 
     Generally, the accuracy of commercial TDR-based material level sensors is on the order of 1%. In order to improve accuracy, the TDR timing system would need a stability on the order of a few picoseconds over time and temperature. Thus, a very precise pulse detection and timing system is needed that is not available in the prior art. 
     SUMMARY OF THE INVENTION 
     The present invention is a time domain reflectometer (TDR) having a single wire transmission line which is inserted into a tank or container, wherein the round trip travel time of reflected pulses indicates the location or, equivalently, the fill-level of the tank. Accurate measurements are made by measuring the difference in reflection times between a reflection at the top of the tank (designated T herein) and a reflection from the material in the tank (designated M herein). This T−M time difference is independent of interconnect cable lengths and propagation delays in the TDR apparatus. Consequently, accurate, stable measurements are possible at the picosecond level. The present invention is also a number of individual components used in the TDR. 
     In order to launch a pulse onto a single wire transmission line, a pulse launcher is needed, such as a coaxial horn or a well-grounded metal plate as used in the prior art. The present invention advantageously employs a sparse, open horn formed of several wires or leaves in place of the prior art pulse launchers to (1) provide a sharp, controlled-impedance discontinuity and thus a sharp, controlled-amplitude reflection, (2) efficiently launch a pulse onto the line, and (3) provide a smooth transition from the horn to free space to avoid spurious reflections at the horn rim. 
     An efficient pulse launcher, as provided by the present invention, virtually eliminates a hot ground effect commonly seen with plate-type launchers. With the open-wire horn, TDR pulses are partially reflected back to the TDR apparatus and partially transmitted onto the dipstick, and very little propagates backwards over the outside of the wire horn launcher and onto the outer jacket of the feed coaxial wire. Were this to occur, ringing and spurious reflections can usually be observed in combination with the desired reflections, making accurate measurements impossible. 
     Mechanically, the wire horn is simple, robust, and inexpensive. Notably, its wires can be bent inwards, in a similar fashion to folding an umbrella, so it can be inserted through a small tank opening such as a ½″ threaded pipe opening. This feature greatly expands the range of applications for the present invention, such as for monitoring the oil level in standard 200 gallon heating oil tanks used throughout the northern U.S., which are commonly fitted with several top-side pipe-threaded openings. 
     In the present invention, a squarewave pulse is transmitted by the TDR apparatus and the return reflections are differentiated into impulses and subsequently sampled to produce an equivalent time (ET) video signal that is an exact replica of the realtime pulses, except on a vastly expanded time scale. Equivalent time techniques convert nanosecond events to millisecond events for vastly simplified processing. 
     The present invention includes a novel low aberration TDR pulse generator having one sharp edge used for measurement, and one slow, return-to-zero edge that has no effect on the system. In addition, a novel TDR circuit is employed to convert the transmitted TDR squarewaves to sharp impulses for accurate, time-of-peak measurement. As a further feature, a novel 2-diode sampler with extremely low line loading and blowby is utilized. 
     Amplitude-gated time-of-peak (TOP) detectors are employed to accurately detect reflected pulses and trigger timing counters. The TOP detectors are independent of pulse amplitude, and are accordingly independent of material dielectric constants, pulse risetime, pulse amplitude, manufacturing variations, long-term drift, and low frequency ringing. 
     In one embodiment, the accuracy of the system is further improved with a unique two-frequency, crystal-controlled timing system that yields scale-factor stabilities limited by the accuracy of a quartz crystal, which is typically ˜0.001%. Alternatively, the quartz crystal may be replaced with a temperature compensated crystal oscillator (TCXO), an ovenized crystal oscillator, or an atomic clock, all of which can provide stabilities well below 1 ppm/° C. 
     The present invention can be used as an electronic dipstick for innumerable applications in material level sensing in containers. In combination with a valve, it can be used to control or automatically regulate the level in a toilet tank, for example. In a totally different application, it can sense the presence and location of an object in contact (or near contact) with its Goubau line, such as a security wire around a window. As a linear displacement transducer, where a moveable reflecting object slides along the Goubau line, vehicle height can be sensed or hydraulic cylinder displacement can be measured for safety or automatic control. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a diagram of a single wire tank level sensor with a wire launch horn of the present invention. 
     FIG. 2 a  depicts the wire horn and associated transmission lines of the present invention. 
     FIG. 2 b  is a plot of impedance versus distance for a wire horn and single wire line. 
     FIG. 3 a  shows forward and reflected step pulses and, alternatively, impulses, on a single wire line with a wire horn and with a reflector such as a material or a movable component. 
     FIG. 3 b  is a reflection waveform to a stepped pulse for the configuration of FIG. 3 a.    
     FIG. 3 c  is a differentiated version of FIG. 3 b,  as would occur with impulses or with a differentiator in the TDR system. 
     FIG. 4 is a block diagram of a TDR system of the present invention. 
     FIG. 5 a  is a schematic diagram of a low aberration transmit pulse generator. 
     FIG. 5 b  is a waveform generated by the circuit of FIG. 5 a  at 100 ns/DIV. 
     FIG. 5 c  is a waveform generated by the circuit of FIG. 5 a  at 200 ps/DIV. 
     FIG. 6 is a schematic diagram of a high-speed sampler and video amplifier. 
     FIG. 7 a  is a block diagram of a prior art CFD-type automatic pulse detector. 
     FIG. 7 b  is a block diagram of a time-of-peak (TOP) detector of the present invention. 
     FIG. 8 is a schematic diagram of a time-of-peak (TOP) detector. 
     FIG. 9 is a timing diagram of the TDR system of FIG.  4 . 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     A detailed description of the present invention is provided below with reference to the figures. While illustrative component values and circuit parameters are given, other embodiments can be constructed with other component values and circuit parameters. All U.S. Patents and copending U.S. applications cited herein are herein incorporated by reference. 
     The same elements or features have the same numbers or labels in the various figures. Illustrative waveforms are shown at some locations in the system/circuit diagrams. 
     FIG. 1 is an overview of a single wire material level sensor  10  of the present invention. A time domain reflectometer (TDR) transceiver unit  12  transmits pulses down a coaxial cable  14  to an attached launcher horn  16  comprised of several wires forming a horn shape. The launcher horn facilitates the propagation of electromagnetic waves (EM) that propagate down an attached single wire transmission line  18 , also known as a Goubau line, or herein, a dipstick. When the propagating pulses encounter a material  20 , some or all of the pulse energy is reflected back up the dipstick and into the TDR unit  12 . The TDR unit  12  processes the round trip time to provide a range or fill-level indication  24 . The launcher horn  16  is configured to provide a pulse reflection marking the top of the tank  22 , labeled T. Accordingly, the TDR unit  12  measures the difference in reflection time between a reflection at T and reflection at M, the material level. The time difference T−M is independent of propagation delays along coaxial cable  14  or apparatus delays within the TDR unit  12 . Thus, an accurate fill-level can be obtained with a timing stability limited, in principle, solely by the mechanical stability of the tank. 
     FIG. 2 a  depicts the entire dipstick assembly  30 , which is comprised of a coaxial cable  32  to interconnect a TDR unit  34  to dipstick rod or wire  36 . A pulse launcher  40  is comprised of wires  38  arranged symmetrically about the dipstick rod or wire  36  and extending outwardly to approximate a horn shape. The horn wires are supported by a small metal plate  39  which is in metallic (i.e. electrical) contact with the shield of cable  32  and the horn wires  38 . 
     FIG. 2 b  plots impedance Z versus distance D and has a one-to-one distance correspondence with FIG. 2 a  along the horizontal axis. The following description relates to both FIGS. 2 a  and  2   b.  Coaxial cable  32  has a nominal impedance of 50Ω. There is a sharp impedance discontinuity  44  at point T (see FIG. 2 b ). (The impedance then ncreases to 500Ω.) The mounting location  42  of the horn wires  38  relative to the dipstick rod or wire  36  defines the impedance at location T, which is a reference location designating the top of the tank. Thus the magnitude of discontinuity  44  can be scaled as a matter of design choice. For very small discontinuities, such as a transition from 50Ω to 60Ω, for example, it may be necessary to broaden the wires into a triangular shape (or open leaf)  46 . In principle, any number of wires may be used, but four wires are preferred. 
     Impedance Z versus distance D smoothly increases as seen at curve  50  In FIG. 2 b.  This smoothness is needed to avoid any potential confusion with a discontinuity produced by a low dielectric constant material. As shown in FIG. 2 a,  wires  38  can have a flare  48  at the ends (even bending back 180°) to improve the smoothness of the transition to the high (500Ω) impedance of the line  36 . Experiments show that the exact angle  52  that the wires make to the dipstick rod are not critical, nor is the precise shape of the flare  48 . The length of the wires  38  should be several times the effective physical length of the pulse being propagated. 
     If wires  38  are made of a flexible material, e.g., brass, steel, or metallized plastic, they can be temporarily bent inwards as indicated by arrow  54  during installation to enable use with a small tank opening. 
     FIG. 3 a  shows the dipstick  60  in combination with a moveable target (or reflector)  68 . Launcher horn  62 , connected to coaxial cable  32 , launches either step-like pulses  64  or impulses  66  down line  67  towards a reflector (target)  68  that is movable and generally represents a material level M or a point of contact in a linear displacement transducer application. Step-like pulses  70  or impulses  72  are reflected from the reflector (target)  68  and appear inverted since the material is always a lower impedance than the free-space between the horn and the material. 
     FIG. 3 b  is a reflection plot of an embodiment of FIG. 3 a.  The vertical scale is expressed in milli-rho, a measure of the reflection coefficient to a step-like pulse propagating along cable  32 . A sharp rise in reflection is seen at point  44  (location T) that smoothly tapers along curve  50  to about 850 milli-rho, representing an impedance Z of ˜500Ω. A reflection  74  is seen at location M due to the presence of reflector  68 . 
     FIG. 3 c  is a differentiated version of the waveform of FIG. 3 b.  Approximately equal amplitude pulses  76 ,  78  can be seen at locations T and M, respectively. As will be discussed shortly, the time of peak of these pulses will be detected and the difference in their occurrence times will be used as a measure of position M relative to T. 
     A spurious pulse  80  was added to the data plotted in FIG. 3 c  to indicate an aberration that might be produced by a solid horn launcher as seen in the prior art, such as in the aforementioned Ross apparatus, whereby the rim of the horn exhibits a sharp impedance discontinuity with free-space. A solid horn is not sufficiently airy compared to a wire horn, and a smooth transition to free-space is nearly impossible. The resulting spurious pulse  80  may false trigger the TDR or create a very large measurement error when M is moved closer to T so its reflection coincides with pulse  80 . 
     FIG. 4 is a diagram of a TDR system  90  of the present invention. To simplify signal processing and to make the entire system practical, the present invention employs expanded time techniques, also known as equivalent time (ET). ET is a beat-frequency effect produced by sampling reflections at a slightly slower rate than the transmitted pulse rate. The net effect is very similar to shining a strobe light on a fan blade, and adjusting the strobe frequency so the blade appears to rotate very slowly. By this analogy, the rapidly rotating fan blade represents the realtime pulses travelling at the speed of light, the strobe is an electronic gate in the receiver (or the gated sampler described herein), and the slowly rotating visual effect is the expanded time millisecond-scale video output  107 . “Video” is used here in the common radar parlance, and is not to be confused with television or visual signals. 
     An ET pulse-echo TDR system transmits pulses, and after a delay its receiver, i.e., its sampler, is gated at a particular point in time, or equivalently, in range. The timing of the gate is typically swept across a range of delays (e.g., 0-100 ns) in a matter of milliseconds, such that the sampler video output is a scan-like waveform which replicates events occurring on a realtime 0-100 ns scale on an equivalent time millisecond-scale. Equivalent time techniques are commonly used in wideband sampling oscilloscopes and will not be dwelt upon here. 
     TX CLOCK  92  in FIG. 4, e.g. a first crystal oscillator, typically operates at 4-megaHertz and triggers pulse generator  94  to produce a squarewave with a fast edge. Optionally, the TX CLOCK can be noise modulated in frequency by noise source  120  to spread the spectrum of the small amount of leakage radiation from the dipstick. The TDR squarewave passes through attenuator resistor  96  and therefrom propagates down microstrip  98  to the dipstick via coaxial cable  32 . Optionally, cable  32  and/or microstrip  98  can be omitted as a design choice, i.e. either cable  32  or the dipstick itself can be directly connected to resistor  96 . 
     Reflections from the dipstick pass through differentiation network (differentiator)  100  to high-speed sampler  104 , which is gated by pulse generator  116  with controlled timing. The sampler output is amplified by amplifier  106  of gain -A to produce a video output signal  107  which is processed by processor  108  to produce a reflection range indication signal  110 . 
     Reflections returning from the dipstick generally must be well-terminated to prevent unwanted triple-transit reflections or pulse “rattles”. Since resistor  96  is typically 470Ω, and since the differentiation network  100  is typically comprised of a 56Ω resistor and a 1 pF capacitor, a true 50Ω termination does not exist by virtue of their combination. Thus, reactive termination network  102  with a 56Ω resistor and a 3 nH inductor has been added to form a real, i.e., non-reactive, 50Ω termination. In order to maintain a non-reactive 50Ω termination, the time constants of networks  100  and  102  must be the same, or about 56 ps in this example. 
     RX CLOCK  114  is typically a second crystal oscillator set to 4 MHz-Δ, where 4 MHz is the frequency of the first oscillator (TX CLOCK) and where A is a small offset, typically 25 Hz, from the TX CLOCK. Thus, the RX CLOCK smoothly slips 360° in phase 25 times per second and thus produces a slow, linear time scan of the reflection pulses present at sampler  104 . A control circuit  112  compares the frequency Δ of the range signal on line  110  (typically a PWM pulse) to a reference frequency Δ ref  and controls RX CLOCK  114  to maintain a precise 25 Hz offset from 4 MHz. Alternatively, control circuit  112  may directly compare the TX and RX CLOCKs via line  111  to regulate the offset Δ. This method is less-preferred since it introduces the TX CLOCK to the RX CLOCK side of the system, raising the possibility of phase contamination and subsequent nonlinear phase slippage. 
     In another clock architecture, the RX CLOCK may operate with a crystal operating at a harmonic of the TX CLOCK plus a small offset Δ, and when combined with a pulse selector circuit, a limited-range sweep can be obtained, such as from 0 to 36°. Dual crystal timing systems are described in co-pending application “Self Locking Dual Frequency Clock System”, Ser. No. 09/282,947, by McEwan, and “Precision Radar Timebase Using Harmonically Related Offset Oscillators”, U.S. Pat. No. 6,072,427, by McEwan. 
     Yet another timing method is obtained by disconnecting and removing the RX CLOCK, as indicated by “X”  119 , and installing a swept timing system, as indicated by connecting the line passing through “X”  121 . In this case timing circuit  118  is swept across a range of delays by a sweep input, typically an analog voltage ramp. The timing sweep usually repeats at a 25 Hz rate and sweeps over a 0-100 ns delay relative to the TX CLOCK to produce an. equivalent time video signal of the reflected pulses. Swept timing circuits having scale factor accuracies on the order of several tens of picoseconds or better can be realized with a Delay Locked Loop (DLL) such as a “Precision Digital Pulse Phase Generator” as disclosed by McEwan in U.S. Pat. No. 5,563,605, or in copending application, “Phase-Comparator-Less Delay Locked Loop”, Ser. No. 09/084,541, now U.S. Pat. No. 6,055,287, by McEwan. 
     FIG. 5 a  is a schematic diagram of a low aberration transmit pulse generator  130 . A logic inverter  132  is coupled through a drive network  134  and diode  136  to a switch transistor  138 . When the inverter swings positive the transistor is rapidly biased on and generates a very fast negative-going transition  140 , as seen in FIG. 5 b  and on an expanded scale in FIG. 5 c.  The transition time is 0.1 ns. 
     When the output of inverter  132  swings low, diode  136  ceases to conduct, but transistor  138  continues to conduct for a short period due to a saturation delay. This saturation delay is used advantageously to eliminate a feed-through spike from the logic inverter. In other words, the transistor&#39;s collector remains clamped to ground at  144  in FIG. 5 b  while its base drive swings low. The transistor finally pulls out of saturation at  146  and its collector returns to its high state  149  at a relatively slow rate  142  determined to a large extent by base bias resistor  148  (see FIG. 5 a ). The slow risetime results in very little signal getting past the TDR differentiator  100 . 
     In TDR circuits such as disclosed herein, it is generally important to avoid coupling glitches onto the line since they may distort the reflections from the material being sensed. After differentiation, seemingly minor glitches are greatly magnified; a glitch-free transmit waveform is essential. Accordingly, the circuit of FIG. 5 a  provides one fast edge for reflectometry and a 100× slower edge during return-to-zero. 
     FIG. 6 is a schematic diagram of a high-speed sampler and video amplifier circuit  160 . Diodes  162 ,  164  comprise high speed sampling diodes in a sampling circuit that further includes switching transistor  166 , sampling capacitor  168 , charge transfer resistor  170 , charge holding capacitor  172 , bias resistor  174 , coupling capacitor  176  and op amp  178 . Except for diode  162  and resistor  163 , this circuit has been fully described in co-pending application “Charge Transfer Wideband Sample-Hold Circuit” Ser. No. 09/084,502, now U.S. Pat. No. 6,060,915, by McEwan, and will not be expanded upon here. 
     The series combination of diodes  162  and  164  results in one-half the input capacitance of a single diode sampling circuit and thus presents less reactive loading to the TDR line and therefore a better termination for reflected pulses. Resistor  163  is situated between diodes  162 ,  164  to shunt glitches from blowing by transistor  166  through capacitor  168  and diodes  162 ,  164  and onto the dipstick. These blowby components are generally in the form of digital logic glitches that can couple through the diodes  162 ,  164  and appear on the dipstick line as spurious pulses. Hence, resistor  163  provides a shunt path to ground to substantially reduce this effect. 
     FIG. 7 a  is a block diagram of a prior art constant fraction discriminator (CFD) automatic pulse detector  180 . Positive peak detector  182  and negative peak detector  184  detect the peak values of repetitive TDR pulses  181  (only one repetition of a 25 Hertz repetition rate is shown for clarity). The outputs of the peak detectors are multiplied by a constant (˜0.5) via voltage divider networks  186 ,  188  respectively, and are then applied as threshold levels to respective comparators  190 ,  192 . Whenever the input pulse amplitude, which is applied to the other input of each comparator  190 ,  192 , exceeds either the positive or negative threshold, the respective comparator triggers a flip-flop  194  to generate a range PWM (pulse width modulation) pulse. The width of the PWM pulse varies with the position of edge  196  and indicates the time difference between the T and the M reflections, or equivalently the fill level of the tank. The scale factor of the PWM pulse is typically 1 us=1 mm. Most importantly, if the pulse amplitude doubles, so will the peak detector outputs and their corresponding threshold levels. Consequently, the exact trigger points,  185 ,  187 , as a percentage of peak amplitude, remain constant. 
     The CFD circuit works well in tracking pulse amplitude variations under ideal conditions. If the pulse amplitude varies rapidly, as may be the case with sloshing materials, the peak detectors will not track rapid decreases in amplitude, since they generally have a fast attack and a slow decay characteristic (˜1-second time constant). Thus, the CFD is unsuitable for all mobile applications, including automotive, aircraft and marine use. A further problem with the CFD is low frequency aberrations such as ringing and baseline tilt that can cause substantial errors. Finally, the CFD of FIG. 7 a  is incomplete; some means of thresholding is needed for the case when there is no reflected pulse and the CFD false triggers on baseline noise. 
     FIG. 7 b  is a block diagram of a time-of-peak (TOP) detector  200  of the present invention. A TOP detector generally detects the occurrence of the peak of a pulse by differentiating the pulse and detecting when a zero axis crossing occurs. Thus the TOP detector  200  has a differentiator  204  coupled through logic gates  208 ,  210  to flip-flop  212  to generate a PWM high level  214  when the slope of the input pulse first goes negative at T or point  203  on the video waveform, and then the PWM pulse flips low at M (edge  216 ) on the PWM pulse or corresponding point  205  on the video waveform. The width of the PWM pulse thereby indicates the T−M time difference, or equivalently the tank fill level. 
     In the absence of input pulses, differentiator  204  generates a high level of random noise at its output. Positive and negative threshold detectors  202  and  206  have been added to eliminate this problem. The output of the threshold detectors change state whenever the input pulses  203 ,  205  exceed predetermined levels (incorporated within the functional blocks), and enable respective gates  208 ,  210 . One gate is shown as an AND gate  208  and the other as an OR gate  210  to suit the specific logic requirements of flip-flip  212 . 
     For any pulse above threshold, its time-of-peak detection is independent of pulse amplitude. TOP detection is also very much independent of baseline tilt, as evidenced, for example, at point  215  of FIG. 3 c.  Unlike the CFD, the TOP detector has no analog memory and can respond to rapid changes in pulse amplitude; it is ideal for sloshing liquids. 
     FIG. 8 is a schematic diagram of the time-of-peak (TOP) detector  220  of FIG. 7 b.  Positive and negative threshold detectors  222 ,  226  and differentiator  224  are based on op amps, and logic AND and OR functions are performed by diode pairs  228 , and  230  respectively. Flip-flop  232  is a D-input type 74HC74. The op amps are TI type TLO74 and the diodes are 1N4148s. The inputs +Vth and −Vth are the threshold voltages applied to threshold detectors  222 ,  226  respectively. 
     FIG. 9 is a timing diagram of the TDR system of FIG.  4  and the TOP detector of FIG. 7 b.  Equivalent time TDR pulses are labeled VIDEO and generate positive and negative threshold pulses labeled POSITIVE GATE and NEGATIVE GATE via threshold detectors  202 ,  206 . The differentiator  204  generates the DIFFERENTIATOR-bar waveform. These waveforms are gated by gates  208 ,  210  to produce POSITIVE GATED DIFFERENTIATOR and NEGATIVE GATED DIFFERENTIATOR pulses that toggle flip-flip  212  to produce the RANGE PWM OUT pulse, indicating material fill level. The circles on the waveforms indicate the TOP trigger points. Note that the gate waveforms simply enable the DIFFERENTIATOR-bar pulses but have no impact on the actual timing measurement. 
     Although the invention has been described with reference to a single wire (or Goubau line) dipstick, the principles of the timing circuitry, the TDR reactive termination and differentiator, the sampler and the TOP detector apply to other TDR embodiments as well, such as a coaxial line and 2-wire dipstick. 
     Changes and modifications in the specifically described embodiments can be carried out without departing from the scope of the invention which is intended to be limited only by the scope of the appended claims.