Abstract:
A passive mixer includes a transconductance amplifier having a source degeneration capacitance. The transconductance amplifier has an input for receiving an input signal and an output for outputting a current signal. A multiplier is provided for mixing a local oscillator signal with the current signal so as to provide an output signal at an output of the passive mixer. A capacitive load is connected to the output of the passive mixer.

Description:
FIELD OF THE INVENTION 
       [0001]    The present invention relates to circuits generally, and more specifically to passive mixers. 
       BACKGROUND 
       [0002]    Capacitor banks tuned by calibration circuitry can be applied for process-voltage-temperature (PVT) compensation in passive mixers.  FIG. 1  shows the circuit of a double-balanced current-mode passive mixer  100 , loaded with tunable capacitor banks C 1 , C 2 , . . . , C N . First, the input voltage V in  is converted to current I in  by a transconductance amplifier G m . Then the current I in  is fed into the switches controlled by complimentary local oscillator (LO) signals, and is accumulated on the load capacitor banks C 1 , C 2 , . . . , C N . 
         [0003]    The local oscillator (LO) generates complimentary signals which are beat against the signal of interest to mix it to a different frequency. The local oscillator (LO) signals control the switches in mixer  100 , while the signal of interest is injected at the input of the mixer  100 , to produce the sum and difference of their frequencies. These are the beat frequencies. Normally for a down converter, the beat frequency is the difference between the two; while for an up converter, the beat frequency is the sum of the two. 
         [0004]    The current accumulated on the capacitor banks C 1 , C 2 , . . . , C N  provides the output voltage V out . The capacitor banks C 1 , C 2 , . . . , C N  are controlled by a calibration circuit  102 , providing a total capacitance of: 
         [0000]    
       
         
           
             
               
                 
                   
                     C 
                     total 
                   
                   = 
                   
                     
                       ∑ 
                       
                         k 
                         = 
                         0 
                       
                       N 
                     
                      
                     
                       
                         h 
                         k 
                       
                        
                       
                         C 
                         k 
                       
                     
                   
                 
               
               
                 
                   ( 
                   1 
                   ) 
                 
               
             
           
         
       
       
         
           
             where h k =0 or 1, representing the digital control bits; and C k =C 1 , C 2 , . . . , C N . 
           
         
       
     
         [0006]    The conversion gain CG of the mixer can be derived as: 
         [0000]    
       
         
           
             
               
                 
                   CG 
                   = 
                   
                     
                       
                         v 
                         out 
                       
                       
                         v 
                         
                           i 
                            
                           
                               
                           
                            
                           n 
                         
                       
                     
                     = 
                     
                       
                         2 
                         π 
                       
                       · 
                       
                         
                           G 
                           m 
                         
                         
                           2 
                            
                           
                               
                           
                            
                           π 
                            
                           
                               
                           
                            
                           
                             f 
                             out 
                           
                            
                           
                             C 
                             total 
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   2 
                   ) 
                 
               
             
           
         
       
     
         [0007]    where f out  is the output frequency; and G m  is the transconductance of a transconductance amplifier, commonly implemented by active devices, as shown in  FIG. 2 . I B  is the DC biasing current. The transconductance G m  is given by: 
         [0000]    
       
         
           
             
               
                 
                   
                     G 
                     m 
                   
                   = 
                   
                     
                       
                         i 
                         o 
                       
                       
                         v 
                         
                           i 
                            
                           
                               
                           
                            
                           n 
                         
                       
                     
                     = 
                     
                       
                         g 
                         m 
                       
                       2 
                     
                   
                 
               
               
                 
                   ( 
                   3 
                   ) 
                 
               
             
           
         
       
     
         [0008]    where g m  is the small-signal transconductance of the transistors M 1  and M 2 . A transconductance amplifier (g m  amplifier) puts out a current proportional to its input voltage. In network analysis the transconductance amplifier is defined as a voltage controlled current source (VCCS). In field effect transistors, transconductance is the change in the drain/source current divided by the change in the gate/drain voltage with a constant drain/source voltage. Typical values of g m  for a small-signal field effect transistor are also 1 to 10 millisiemens. 
         [0009]    Substituting (3) into (2), provides the mixer conversion gain as: 
         [0000]    
       
         
           
             
               
                 
                   CG 
                   = 
                   
                     
                       
                         v 
                         out 
                       
                       
                         v 
                         
                           i 
                            
                           
                               
                           
                            
                           n 
                         
                       
                     
                     = 
                     
                       
                         1 
                         π 
                       
                       · 
                       
                         
                           g 
                           m 
                         
                         
                           2 
                            
                           
                               
                           
                            
                           π 
                            
                           
                               
                           
                            
                           
                             f 
                             out 
                           
                            
                           
                             C 
                             total 
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   4 
                   ) 
                 
               
             
           
         
       
     
         [0010]    In sub-micron processes, the small-signal transconductance g m  can vary in magnitude by 2-3 times due to PVT variation. Thus, to compensate for the PVT-induced gain variation, the load capacitance C total  covers a wide range, resulting in a large silicon area, especially when a low density capacitor (such as a metal capacitor) is used (for example, to achieve good linearity of the mixer). 
         [0011]    Further, the calibration circuit  102  was needed to detect the mixer gain variation, and then to tune the capacitor banks, resulting in significant power consumption. 
         [0012]    The PVT compensation circuitry and calibration circuitry  102  consume high power and a large silicon area, especially when the PVT-induced gain variation is large, as in the case of current-mode passive mixers  100 . 
       SUMMARY OF THE INVENTION 
       [0013]    In some embodiments, a passive mixer comprises a transconductance amplifier having a source degeneration capacitance. The transconductance amplifier has an input for receiving an input voltage signal and an output for outputting a current signal. A multiplier is provided for mixing a local oscillator signal with the current signal so as to provide an output signal at an output of the passive mixer. A capacitive load is connected to the output of the passive mixer. 
         [0014]    In some embodiments, a method comprises receiving an input voltage signal with a transconductance amplifier having a source degeneration capacitance. A current signal is output from the transconductance amplifier. A local oscillator signal is mixed with the current signal to generate an output signal. The output signal is provided at an output of the passive mixer. The output has a capacitive load connected thereto. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0015]      FIG. 1  is a schematic diagram of a conventional passive mixer having calibration circuitry and tunable capacitor banks. 
           [0016]      FIG. 2  is a schematic diagram of a transconductance amplifier suitable for use in the mixer of  FIG. 1 . 
           [0017]      FIG. 3  is a schematic diagram of a passive mixer according to an exemplary embodiment. 
           [0018]      FIG. 4A  is a schematic diagram of a transconductance amplifier suitable for use in the mixer of  FIG. 3 . 
           [0019]      FIG. 4B  is a schematic diagram of a transconductance amplifier suitable for use in the mixer of  FIG. 3 , including an adjustable capacitor. 
           [0020]      FIG. 5  is a graph showing simulated variation in output of the conventional mixer of  FIG. 1  at three different PVT corners. 
           [0021]      FIG. 6  is a graph showing simulated variation in output of the mixer of  FIG. 3  at the three PVT corners used in  FIG. 5 . 
           [0022]      FIG. 7  is a graph showing simulated output of the mixer of  FIG. 3  as a function of load capacitance. 
           [0023]      FIG. 8  is a schematic diagram of a single-balanced passive mixer according to an exemplary embodiment. 
           [0024]      FIG. 9  is a schematic diagram of an exemplary mixer including an operational amplifier with a feedback capacitor as the capacitive load. 
       
    
    
     DETAILED DESCRIPTION 
       [0025]    This description of the exemplary embodiments is intended to be read in connection with the accompanying drawings, which are to be considered part of the entire written description. 
         [0026]    A current-mode passive mixer  300 ,  800 ,  900  with conversion gain that is independent of process, voltage, and temperature (PVT) variation is described below, with reference to  FIGS. 3-9 . Optionally, the conversion gain may be programmable. Using this technique, the conversion gain of the current-mode passive mixer  300 ,  800 ,  900  is proportional to the ratio of the source degeneration capacitance C S  to the load capacitance C L . Thus, although the absolute capacitance and other circuit parameters vary with PVT, the ratio of the source degeneration capacitance C S  to the load capacitance C L  is independent of PVT variation, and so is the mixer conversion gain. In particular, if C S  and C L  are of the same type of construction, then both are assured to vary in the same manner together in response to any PVT variation, so that the ratio thereof stays constant, independent of the PVT variation. 
         [0027]    This technique eliminates the need for extra PVT compensation circuitry, such as the wide-range capacitor banks (e.g., C 1 , C 2 , . . . , C N  shown in  FIG. 1 ) tuned by calibration circuitry (e.g., circuit  102  in  FIG. 1 ). Moreover, the conversion gain of the exemplary mixer  300  can optionally be made programmable, by switching between different load capacitance C L  (or switching between different source-degeneration capacitance C S ), without relying on any gain detecting circuitry. 
         [0028]    Using the exemplary technique, the conversion gain of the current-mode passive mixer is proportional to the ratio of the source degeneration capacitance C S  to the load capacitance C L , which is independent of PVT variation and is (optionally) programmable without relying on any gain detecting circuitry. 
         [0029]      FIG. 3  shows a passive mixer  300 , including a transconductance amplifier G m,sd  with a source degeneration capacitance C S . In the example of  FIG. 3 , the capacitive load C L  does not require a bank of switchable capacitors to compensate for PVT variation, so there is no need for a calibration circuit to control switching of capacitors within capacitor banks. The mixer  300  may be a down-converting mixer, in which case the input voltage V in  is a high or radio frequency signal, and the output voltage, V out  is a low or intermediate frequency signal. Alternatively, the mixer  300  may be an up-converting mixer, in which case the input voltage V in  is a low or intermediate frequency signal, and the output voltage, V out  is a high or radio frequency signal. 
         [0030]    First, the input voltage V in  is converted to current I in  by a transconductance amplifier G m,sd . Then the current I in  is fed into the switches controlled by complimentary local oscillator (LO) signals, and is accumulated on the load capacitance C L . 
         [0031]    The local oscillator (LO) generates complimentary signals which are beat against the input signal V ,in  to mix it to a different frequency. The local oscillator (LO) signals control the switches in mixer  300 , while the signal V ,in  is injected at the input of the mixer  300 , to produce the sum and difference of their frequencies, one of which (depending on the goal of mixing, either up-conversion or down-conversion t ) will be the output frequency of interest. 
         [0032]      FIG. 4A  is a schematic diagram of the transconductance amplifier G m,sd  shown in  FIG. 3 .  FIG. 4A  introduces a capacitive source degeneration structure Cs to the transconductance amplifier G m,sd , shown in  FIG. 3 . The transconductance is given by: 
         [0000]    
       
         
           
             
               
                 
                   
                     
                       G 
                       
                         m 
                         , 
                         sd 
                       
                     
                     = 
                     
                       
                         
                           i 
                           o 
                         
                         
                           v 
                           
                             i 
                              
                             
                                 
                             
                              
                             n 
                           
                         
                       
                       = 
                       
                         
                           
                             1 
                             2 
                           
                           · 
                           
                             
                               g 
                               m 
                             
                             
                               1 
                               + 
                               
                                 
                                   g 
                                   m 
                                 
                                 
                                   2 
                                    
                                   
                                       
                                   
                                    
                                   π 
                                    
                                   
                                       
                                   
                                    
                                   
                                     f 
                                     
                                       i 
                                        
                                       
                                           
                                       
                                        
                                       n 
                                     
                                   
                                    
                                   
                                     C 
                                     S 
                                   
                                 
                               
                             
                           
                         
                         ≈ 
                         
                           π 
                            
                           
                               
                           
                            
                           
                             f 
                             
                               i 
                                
                               
                                   
                               
                                
                               n 
                             
                           
                            
                           
                             C 
                             S 
                           
                         
                       
                     
                   
                   , 
                   
                     
 
                   
                    
                   
                     
                       when 
                        
                       
                           
                       
                        
                       
                         
                           g 
                           m 
                         
                         
                           2 
                            
                           π 
                            
                           
                               
                           
                            
                           
                             f 
                             
                               i 
                                
                               
                                   
                               
                                
                               n 
                             
                           
                            
                           
                             C 
                             S 
                           
                         
                       
                     
                     &gt;&gt; 
                     1 
                   
                 
               
               
                 
                   ( 
                   5 
                   ) 
                 
               
             
           
         
       
     
         [0033]    where f in  is the frequency of the input signal V in . 
         [0034]    With this transconductance amplifier G m,sd , a current-mode passive mixer  300  loaded with a single load capacitance C L  is implemented, as shown in  FIG. 3 . The conversion gain of the mixer  300  is provided by the equation: 
         [0000]    
       
         
           
             
               
                 
                   
                     
                       
                         CGsd 
                         = 
                           
                          
                         
                           
                             
                               v 
                               out 
                             
                             
                               v 
                               
                                 
                                     
                                 
                                  
                                 
                                   i 
                                    
                                   
                                       
                                   
                                    
                                   n 
                                 
                               
                             
                           
                           = 
                           
                             
                               2 
                               π 
                             
                             · 
                             
                               
                                 G 
                                 
                                   m 
                                   , 
                                   sd 
                                 
                               
                               
                                 2 
                                  
                                 π 
                                  
                                 
                                     
                                 
                                  
                                 
                                   f 
                                   out 
                                 
                                  
                                 
                                   C 
                                   L 
                                 
                               
                             
                           
                         
                       
                     
                   
                   
                     
                       
                         = 
                           
                          
                         
                           
                             2 
                             π 
                           
                           · 
                           
                             
                               π 
                                
                               
                                   
                               
                                
                               
                                 f 
                                 
                                   i 
                                    
                                   
                                       
                                   
                                    
                                   n 
                                 
                               
                                
                               
                                 C 
                                 S 
                               
                             
                             
                               2 
                                
                               π 
                                
                               
                                   
                               
                                
                               
                                 f 
                                 out 
                               
                                
                               
                                 C 
                                 L 
                               
                             
                           
                         
                       
                     
                   
                   
                     
                       
                         = 
                           
                          
                         
                           
                             1 
                             π 
                           
                           · 
                           
                             
                               
                                 f 
                                 
                                   i 
                                    
                                   
                                       
                                   
                                    
                                   n 
                                 
                               
                                
                               
                                 C 
                                 S 
                               
                             
                             
                               
                                 f 
                                 out 
                               
                                
                               
                                 C 
                                 L 
                               
                             
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   6 
                   ) 
                 
               
             
           
         
       
     
         [0035]    In equation (6), the frequencies f in , and f out  are generally pre-defined in a given system. Thus the conversion gain of the mixer is proportional to the ratio of the source degeneration capacitance C S  and the load capacitance C L . If Capacitors C S  and C L  are the same type of capacitors, the ratio of the capacitance C S /C L  is independent of PVT variation, and so is the mixer conversion gain CG sd . Thus, extra PVT compensation with capacitor banks tuned by extra calibration circuitry is not necessary, saving both power and silicon area. 
         [0036]    In some embodiments, if the conversion gain CG sd  of the mixer  300  is desired to be programmable, the capacitive load C L  may be provided by variable capacitors, or the source degeneration capacitors C S  may be variable capacitors. Choosing different values of the load capacitor C L  (or choosing different values of the source degeneration capacitor C S ), the conversion gain is precisely determined by the ratio C S /C L . Gain detection circuitry is not necessary. 
         [0037]      FIG. 4B  shows a variation of the transconductance amplifier of  FIG. 4A , in which the source degeneration capacitance is optionally provided by a controlled capacitor in either the analog fashion or the digital fashion. The remaining elements of  FIG. 4B  are the same as in the transconductance amplifier of  FIG. 4A . 
         [0038]      FIGS. 5 and 6  show simulation results comparing the output voltages of the conventional structure (shown in  FIG. 1  and  FIG. 2 ) and the exemplary structure (shown in  FIG. 3  and  FIG. 4 ), applying the same input signal. Both structures are down-converting the input high-frequency signal to an output low-frequency signal. The output frequency is the difference of the input frequency and the local oscillator (LO) frequency. Different PVT corners were applied, which induced variation of g m , C total  (without any tuning), C L , and C S . As shown in  FIG. 5 , the output of the conventional structure of  FIGS. 1 and 2  varies around 6 dB from the slowest PVT corner to the fastest PVT corner.  FIG. 6  shows that the output of the exemplary structure of  FIGS. 3 and 4  varies only 0.6 dB from the slowest PVT corner to the fastest PVT corner. 
         [0039]      FIG. 7  shows the programmability of the exemplary structure according to simulation results. By switching between different values of C L  (10 pF, 20 pF, and 40 pF), the conversion gain of the mixer varies accordingly. The peak amplitudes in  FIG. 7  are as predicted by equation (6). As noted above, the programming may be adjustable, through use of an adjustable capacitor C S  or C L  or the programming may be implemented by selecting a fixed source degeneration value from among various capacitance values. 
         [0040]    Using the exemplary technique, the conversion gain CG sd  of the current-mode passive mixer  300  is proportional to the ratio of the source degeneration capacitance C S  to the load capacitor C L . Thus, the conversion gain CG sd  is programmable and is independent of PVT variation. Compared to the use of capacitor banks C 1 , C 2 , . . . , C N  tuned by extra calibration circuitry  102  for PVT compensation, the exemplary technique consumes less power and less silicon area, and provides programmability without relying on any gain detecting circuitry. 
         [0041]    As noted above, the mixer  300  may be a down-converting or up-converting mixer, with the conversion gain of 1/π*(f in *Cs)/(f out *C L ). For a down-converting mixer, since the input frequency f in  is higher than the output frequency f out , it is not difficult to achieve usable gain. For a up-converting mixer where the input frequency f in  is lower than the output frequency f out , to achieve usable gain of the mixer, it is preferred to use the up-converting mixer with high input frequency (such as in a high-IF architectures), or in embodiments in which Cs or C L  (or both of them) are implemented off-chip. 
         [0042]    In  FIGS. 3 ,  4 A, and  4 B, double-balanced and differential configurations are used. Although  FIG. 3  shows a double-balanced structure, the invention is not limited thereto. 
         [0043]      FIG. 8  shows a single-balanced mixer structure  800 , including a single-ended transconductance amplifier  802  with a source degeneration capacitance Cs. Like elements in  FIGS. 3 ,  4 , and  8  are shown by like reference numerals. 
         [0044]    Although in  FIGS. 3 and 8 , a single capacitor is drawn to represent the capacitive load C L , the invention is not limited thereto. Rather, a capacitive load can be any simple or complicated structure which loads the mixer capacitively. 
         [0045]      FIG. 9  shows a double-balanced mixer structure  900 , loaded with an operational-amplifier (OPAMP) with a capacitor in the feedback path. Like elements in  FIGS. 3 ,  8 , and  9  are shown by like reference numerals. 
         [0046]    Although the invention has been described in terms of exemplary embodiments, it is not limited thereto. Rather, the appended claims should be construed broadly, to include other variants and embodiments of the invention, which may be made by those skilled in the art without departing from the scope and range of equivalents of the invention.