Abstract:
A power amplifier includes an input terminal, and a gain control circuit connected to the input terminal. The gain control circuit includes a circuit for providing a control signal, and at least a first pair of transistors. Each transistor includes a control terminal, a first conduction terminal and a second conduction terminal. The control terminals are connected together for receiving the control signal for control thereof. The first conduction terminals are connected together and to the input terminal. The at least one first pair of transistors defines a voltage/current transconductor circuit for converting an input voltage into an input current, and defines a shunt circuit for shunting at least a portion of the input current.

Description:
FIELD OF THE INVENTION 
     The present invention relates to electronics, and, more particularly, to a controlled-gain power amplifier device which is also known as a controlled-gain power driver. The present invention applies advantageously, but is not limited to radio-frequency circuits, particularly circuits used in cellular mobile telephones. 
     BACKGROUND OF THE INVENTION 
     Radio-frequency circuits used in cellular mobile telephones do not incorporate the power amplifier. The power amplifier is generally an external circuit formed using gallium arsenide (GaAs) technology. On the transmission side, the last stage incorporated into the radio-frequency circuit is a power amplifier device. The power amplifier device has a controlled gain, and is also known as a power driver. A power driver typically delivers a maximum level of only a few dBm (0 to 6 dBm) at 1 or 2 GHz depending on the frequency band. The power level delivered by the specific power amplification circuit that is formed within the radio-frequency circuits provides about 27 to 33 dBm. The power amplification circuit is formed from gallium arsenide, for example. 
     Moreover, for new generation telephones using the Code Division Multiple Access (CDMA) mode which includes embedding several tens of communications within the same frequency band (e.g., 1.25 MHz), the power amplification end stage must be furnished with variable-gain control whose dynamic range may reach 20 dB. It is necessary to control the power transmitted in view of a particular average power level requirement in transmissions using the CDMA mode. 
     Finally, the power driver must exhibit linearity constraints, i.e., a linearity of the transfer function linking the output power to the level of the input signal. Stated otherwise, if the input signal level increases linearly, the power output must also increase linearly. The requirements for this last stage of the radio-frequency circuit depend on a compromise between current consumption, linearity, supply voltage, circuit area (influencing the cost of the silicon), and noise floor. 
     Such a device or controlled-gain power amplifier stage receives a signal, which usually originates from an external filter whose output impedance requires a 50 ohm matching for template compliance constraints. One therefore generally finds in series from the input to the output of a power amplifier device an input impedance matching network, a voltage/current transconductor block, gain control carried out by shunting a variable proportion of the current from the transconductor block to the output load, and a network for matching power to the input of the specific power amplification circuit (which follows the integrated radio-frequency circuit). 
     The design most frequently encountered uses an input transconductor block which is a common emitter transistor (single-input version) or two transistors with linked common emitters (differential-input version). These are known for their high power gain. After the input transconductor block is a pair of transistors with linked emitters, which shunt part of the current originating from the transconductor block. The transistors of the transconductor block may be conventionally biased by a decoupled current source. 
     In this type of structure, the linked emitters of the transistors of the current shunting means or circuit are linked to the collector of the transistors of the transconductor block, and are consequently current-driven. Moreover, the input of the device is linked to the bases of the transistors of the transconductor block. Finally, the input matching of this type of stage involves using an inductance connected between the emitters of the transistors of the transconductor block and ground (differential-input version) to form an impedance having a real part with respect to the input. 
     This type of controlled-gain power amplification stage exhibits numerous drawbacks and limitations. First, the greater the linearity requirement, the greater the consumption of the stage. Moreover, such a prior art structure leads to a limitation in the maximum power output. This is because the limitation in the maximum voltage swing can be applied to the output without saturating the output transistor, i.e., the transistors of the current shunting circuit. 
     Also, the conventional bias circuits merely intensify the above limitation by adding the breakdown voltage of the transistor forming the bias current source. Furthermore, whereas the use of an inductance connected between the emitter of the input transistor and ground (single-input version) makes it possible to form an input impedance whose real part is significant, the natural input impedance of a bipolar transistor degenerated by such an inductance depends on numerous parameters. 
     Hence, the input matching network reflects towards the input of the arrangement the drifting of the nominal input impedance of the transconductor block related to the process and temperature variations. This often leads to unstable optimizations which may impair the efficiency of production. It may also result in different characteristics of the various batches produced. 
     Moreover, the order of magnitude of the natural input impedance of a common emitter arrangement is greater than 200 ohms. A 50 ohm matching therefore causes, on crossing the input impedance matching network, an over voltage whose coefficient is in the ratio of the square root of the impedances. The matching is frequently rendered impossible if this coefficient is too large. The matching is then done by degrading its quality coefficient, i.e., by introducing losses which degrade the noise factor. 
     Moreover, the search for high linearity is frustrated by the fact that the rise in the degeneracy inductance results in an increase in the input impedance of the transconductor block, and hence in the voltage at the input of the device. This goes against the sought-after effect since the input transconductor is voltage-controlled. Furthermore, the use of a degeneracy inductance across the terminals of the transistors of the transconductor block is expensive in terms of silicon area. 
     SUMMARY OF THE INVENTION 
     In view of the foregoing background, it is therefore an object of the present invention to provide a controlled-gain power amplifier device offering a better linearity/current consumption compromise, as well as better control of the input impedance of the device. This is applicable to a single input structure and a differential input structure. 
     Another object of the present invention is to provide a saving in terms of silicon area relative to the prior art devices. 
     Yet another object of the present invention is to provide a stable input matching that is easy to achieve. 
     A further object of the present invention is also to provide an arrangement allowing separate optimization of gain, noise and linearity. 
     Yet a further object of the present invention is to provide a biasing that allows accurate control of the quiescent currents, and which is not sensitive to the offset voltage of the biased transistors at high current, and is not sensitive to differences of the coefficients β of these transistors. There are different values for the coefficients β of the transistors formed in the silicon, although in theory they are identical. 
     These and other objects, advantages, and features in accordance with the present invention are provided by a controlled-gain power amplifier device comprising voltage/current transconductor means or a voltage/current transconductor circuit, and gain control means or a gain control circuit comprising shunting means or a shunting circuit able to shunt to the output of the device, in response to a control signal, all or some of the current delivered by the transconductor circuit. A control circuit delivers the control signal. 
     According to a general characteristic of the invention, the device comprises at least one pair of transistors having linked emitters (bipolar transistors) or sources (field effect transistors), controlled at their base or gate by the control circuit. This pair of transistors make up both the transconductor circuit and the shunting circuit. The emitters or sources of the transistors of the pair are furthermore connected to the input of the device. 
     Stated otherwise, the pair of transistors with linked emitters or sources (a single-input architecture) or else the two pairs of transistors with linked emitters or sources (a differential architecture with differential input) form a single active stage ensuring both the transconductance function and the current shunting function. In contrast, these two functions in the prior art devices were carried out by different circuits. 
     Furthermore, these pairs of transistors are now power-driven rather than current-driven. When power driven, these transistors see an impedance equal to their own input impedance while. When current driven (prior art), the transistors of the transconductor block see an impedance which was much higher than their own input impedance. The linearity/consumption compromise is significantly improved since the device comprises only a single active stage instead of, as in the prior art, two cascaded active stages. 
     Moreover, the input impedance of the active stage depends essentially on the emitter resistance and possesses a negligible imaginary part, thereby rendering the input matching easy and stable. Furthermore, this input impedance is small compared with the 50 ohm input matching generally required. This leads to the obtaining of an over voltage coefficient of less than 1, thus aiding the linearity of the device. The small ratio between the input impedance of the active stage and the 50 ohm impedance allows the device to operate even more in a small signal domain, thereby promoting linearity. Lastly, the absence of degeneracy inductance allows a very substantial reduction in the surface expanse of the device. 
     According to one embodiment of the invention, the device may comprise a feedback resistor connected between the emitters or sources of the transistors of the pair and the input of the device. The use of such a feedback resistor, which results in the addition of a further parameter, allows separate optimization of gain, noise and linearity. 
     Furthermore, this feedback resistor ensures extra linearity. However, it also leads to a decrease in the gain of the device. Hence, it is preferable, for certain applications, for the feedback resistance to be between 0.5 times and 3 times the emitter resistance or source resistance of the transistors of the pair. This leads to a good linearity/gain compromise. 
     According to an advantageous embodiment of the invention, the device comprises a biasing circuit for biasing the transistors of the pair. The biasing circuit comprises a bias resistor connected between the emitters or sources of the transistors of the pair and ground, as well as auxiliary circuit to slave the common mode voltage across the terminals of the bias resistor to a reference value. This is done through an amplifier controlling the common mode voltage. 
     Such an embodiment makes it possible to bias the transistors under a minimum breakdown voltage by a feedback amplifier which controls the common mode current of the structure (low frequency loop). A peak output swing equal to 60% of the supply voltage is then possible with a supply voltage on the order of 2.7 volts and a breakdown voltage of 200 mV. This leads to other applications at higher output power, but also to applications for which the supply voltage is even lower than 2.7 volts. 
     It is also particularly advantageous to provide a resistor or an additional inductance connected between the emitters or sources of the transistors of the pair and the bias resistor. Such an additional load makes it possible to increase the radio-frequency impedance of the biasing circuit. This thereby makes it possible to minimize the radio-frequency signal losses in the biasing circuit. 
     The subject of the invention is also applicable to a cellular mobile telephone comprising a controlled-gain power amplifier device as defined above. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     Other advantages and characteristics of the invention will become apparent on examining non-limiting embodiments, and the appended drawings, in which: 
     FIG. 1 is a block diagram of a radio-frequency circuit comprising at the output stage a controlled-gain power amplifier device according to the present invention; 
     FIG. 2 is a more detailed block diagram of the controlled-gain power amplification device illustrated in FIG. 1; 
     FIG. 3 is a schematic diagram of the controlled-gain power amplification device with a differential input structure according to the present invention; and 
     FIG. 4 is a schematic diagram of the controlled-gain power amplification device with a single input structure according to the present invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     In FIG. 1, the reference CRF designates a radio-frequency circuit preceding a power amplifier circuit APP. The circuit is formed using gallium arsenide technology, for example, and is connected to a transmission antenna AT. The assembly may be incorporated into a cellular mobile telephone TMCL, for example. 
     In FIG. 1, the other conventional elements of a cellular mobile telephone have not been represented to simplify the drawing. The radio-frequency circuit CRF receives a baseband signal on its input terminal BB and delivers on its output terminal BS a radio-frequency signal at 1 or 2 GHz, for example, with a maximum level of a few dBm. The power level at the output of the circuit APP can be on the order of 33 dBm. 
     The radio-frequency circuit conventionally comprises one or more frequency transposition stages or mixers (two are represented in FIG. 1) using local oscillator signals OL 1  and OL 2  to perform the frequency transposition. The last stage of the circuit CRF is a controlled-gain power amplification stage or device ETS. This is the subject of the present invention. 
     As illustrated in FIG. 2, the controlled-gain power amplification device ETS comprises, between its input terminal BE and its output terminal BS, an input impedance matching network RAP 1  followed by a voltage/current transconductor block BTC having a transconductance equal to gm. Next there is a gain control circuit BCG formed by shunting a variable proportion of the current from the transconductor block BTC to the output. Finally, there is an output impedance matching network RAP 2 . 
     Conventionally, the function of the input impedance matching network RAP 1  is to present at the input terminal BE a prescribed input impedance Zi such as 50 ohms, whereas the input impedance ZE of the arrangement as seen downstream of the matching network RAP 1  is different. Likewise, the purpose of the output impedance matching network RAP 2  is to present at the output terminal BS a predetermined impedance Zs, such as 50 ohms. 
     Functionally, the transconductor block BTC converts a voltage present at its input into a current equal to the product of this voltage times the transconductance gm. The block BCG delivers to the matching network RAP 2  all or a fraction of this current. This amount may be between 1/1000 and 1, for example. 
     According to the invention, the functions of transconductance BTC and of the current shunting will be carried out within the same active stage ETA, a structural example of which will be detailed while referring more particularly to FIG.  3 . In this figure, which illustrates a structure of the differential type, the differential input BE is composed of two separate terminals. The same holds for the differential output BS. The active stage ETA is composed of two pairs of NPN bipolar transistors referenced T 1 , T 2  and T 3 , T 4 . 
     The two inputs of the stage ETA are made up of the two linked emitters of the two transistors of each pair. The bases of the two transistors of each pair are linked together by two voltage sources ST equal to Vg/2 and−Vg/2 respectively. Moreover, the two collectors of the transistors T 2  and T 3  belonging to each of the two pairs are linked respectively to the supply voltage Vcc. The two collectors of the other two transistors T 1  and T 4  are linked respectively to the two output terminals BS via the output matching network RAP 2 . 
     The linked emitters of the transistors T 1  and T 2  of the first pair and the linked emitters of the transistors T 3  and T 4  of the second pair are connected respectively to the two input terminals BE by way of two feedback resistors RE and of the input matching network RAP 1 . 
     The current delivered at the output of the active stage ETA is equal to the product of the transconductance gm of this stage times the voltage Vin present downstream of the impedance matching network RAP 1 , i.e., between the terminals BP. Moreover, this voltage Vin is equal to the product of the over voltage coefficient Qv times the voltage Ve present upstream of the network RAP 1 , i.e., between the two input terminals BE. The over voltage coefficient Qv is equal to the square root of the ratio ZE/Zi. Moreover, in their current shunting function, the transistors T 1 , T 2 , T 3  and T 4  are controlled by control circuit made up of the voltage sources ST. 
     More precisely, the difference between the collector current of the transistor T 1 , for example, and the collector current of the transistors T 2  and T 3 , is proportional to the emitter current I 0 , and depends moreover on the control voltage Vg. Also, depending on the value of the control voltage Vg, the collector current of the transistors T 2  and T 3  may be higher or lower. Consequently, all or some of the current delivered by the stage ETA may actually be delivered at the output of the device, i.e., the terminals BS. 
     In the absence of feedback resistance RE, the impedance ZE is substantially equal to an emitter resistance 2re 0  equivalent to the sum, i.e., of the two parallel emitter resistances of the transistors T 1  and T 2 , or the two parallel emitter resistances of the transistors T 3  and T 4 . Therefore, the transconductance gain of the active stage ETA is equal to gm 0 , where gm 0  is proportional to 1/re 0  and depends only on the bias current of these transistors. 
     Apart from the advantage achieved by the presence of a single active stage ETA instead of two cascaded stages in the prior art device, the input impedance ZE is real (zero imaginary part) and perfectly controlled, thus allowing easy and stable input matching. Furthermore, better intrinsic behavior of the drive by the emitter is obtained in view of the very good control of the input impedance and of the small voltage excursion on the emitter since, in the present case, the overvoltage coefficient Qv is less than 1. The conventional values of emitter resistances are on the order of 1 to 3 ohms, while Zi is equal to 50 ohms, for example. 
     The presence of the feedback resistor RE makes it possible to add a parameter which permits separate optimization of the gain, noise and linearity. Thus, with the presence of the resistor RE, the impedance ZE is substantially equal to 2(RE+re 0 ) and the transconductance gm of the stage ETA is then equal to gm 0 /(1+gm 0 *RE). 
     A feedback resistance RE between 0.5 times the emitter resistance re 0  and three times this emitter resistance ensures a good compromise between the decrease in gain and the increase in linearity. A feedback resistance equal to twice the emitter resistance re 0  may be chosen. 
     Although it would be possible to bias the transistors T 1  to T 4  with conventional biasing circuits, it is preferable to obtain a low breakdown voltage of the arrangement using the biasing circuit described in FIG.  3 . More precisely, these comprise two bias resistors R 0 , respectively connected between ground and the emitters of the transistors T 1  and T 2 , or T 3  and T 4 . 
     The biasing circuit also comprises an auxiliary circuit comprising a linear amplifier CMP whose output is linked to the bases of the transistors T 1  to T 4  via the two voltage sources ST. The +input of the amplifier receives a reference voltage Vref, and the inverting input of the amplifier is connected to the midpoint of a resistive bridge R 1 . R 1  is disposed between the bias resistors R 1 . 
     The common mode voltage across the terminals of the bias resistors R 0  is thus slaved to the value Vref through the amplifier CMP controlling this common mode voltage of the transistors T 1  to T 4 . The bias current I 0  when quiescent (i.e., in the absence of any signal at the input) is then constant, predetermined, and is equal to Vref/R 0 . 
     Therefore, taking a voltage Vref equal to 200 mV and a resistance R 0  equal to 10 ohms, a current I 0  of 20 mA and a voltage across the terminals of the resistors R 0  equal to 200 mV are obtained. A low breakdown voltage of the arrangement is therefore obtained. The voltage is reduced to VBE +VO, where VBE designates the base-emitter voltage drop of a bipolar transistor. Thus, under a supply Vcc of 2.7 volts, with VO equal to 200 mV, it is possible to obtain a peak output swing of 60% of the value of Vcc. 
     It is also advantageous to incorporate two additional resistors RA of 10 ohms, for example, between the two resistors R 0  and the emitters of the transistors T 1  and T 2  or T 3  and T 4 . These additional resistors make it possible to increase the radio-frequency impedance of the biasing circuit, and they thus minimize the radio-frequency signal loss in these biasing circuits. 
     The example just described uses a differential structure. However, a structure with a single input, such as that illustrated in FIG. 4, is also possible. In this FIG. 4, elements which are similar or which have functions similar to those illustrated in FIG. 3 are assigned the same references. The manner of operation and the advantages of this embodiment will therefore not be described in detail here, as they are identical to those set forth in conjunction with FIG.  3 . 
     In this single-input embodiment, the active stage ETA comprises only a single pair of transistors T 6  and T 7 , the collector of one of the transistors (T 6 ) is linked to the output terminal BS, whereas the collector of the other transistor (T 7 ) is linked to the supply voltage +Vcc. 
     The invention has been described as using more particularly NPN bipolar transistors. The invention also applies to a device using transistors of opposite polarity via a reversal between the supply voltage and ground. The invention also applies to N-channel or P-channel MOS transistors. The source, the drain and the gate of these MOS transistors are functionally equivalent to the emitters, collectors and bases of the bipolar transistors. 
     Finally, in FIGS. 2 to  3  the input and output matching networks have been represented as forming an integral part of the devices according to the invention. This being so, these matching networks may be entirely within or in part of the device.