Abstract:
A low-noise, linearized double-balanced active mixer circuit is described, including a first input for a local oscillator (LO), a second input for an intermediate frequency (IF) signal, and an output for a resulting product radio frequency (RF) signal. The mixer circuit also includes a feedback transformer circuit for the purpose of improving the intermodulation (IM) performance. The lossless nature of the feedback topology gives the active mixer a lower noise figure (NF) characteristic than is realizable with conventional methods. The number of active devices is minimized in order to ensure that the mixer attains the lowest possible NF.

Description:
BACKGROUND OF THE INVENTION 
     Mixers are used in communications circuits for the purpose of generating a modulated carrier for transmission, demodulating a modulated carrier in reception, or converting a signal at some input intermediate frequency (IF) to another output radio frequency (RF) by multiplying two input signals and thereby generating a third. A number of mixer realizations, both passive and active, are known in the art, and double-balanced mixers are known particularly well due to their advantages in the suppression of unwanted spurious signals and the isolation of any one of three ports to the other two, there generally being two inputs and one output. The Gilbert Cell has been the most widely used active mixer circuit for performing the above tasks, usually incorporated within an integrated circuit. It does, however, possess certain limitations in terms of intermodulation (IM) distortion and noise figure (NF) that precludes it&#39;s use in communications systems having demanding performance specifications. The series-shunt feedback mixer delivers a much improved IM performance, but the lossy nature of the feedback topology does not improve the NF performance. The lossless feedback mixer offers an improvement in noise figure, and this performance can be further improved by a simple modification. 
     Referring to FIG. 1, a schematic diagram of a lossless feedback double-balanced active mixer is shown in functional form. Here, the mixer is comprised of switching transistors  101 ,  102 ,  104 , and  105 , which are turned on (saturation) and off (cutoff) alternately by a differentially applied local oscillator (LO) signal. By this switching action, a pair of currents generated by driver transistors  103  and  106  are divided into four paths, there being two paths for each of two currents. The currents generated by driver transistors  103  and  106  are the result of an input intermediate frequency (IF) signal applied differentially to the input windings of a pair of feedback transformers  107  and  108 . The hybrid transformers  111  and  112  combine the four currents from switching transistors  101 ,  102 ,  104 , and  105 , creating a differential pair of feedback currents  119  and  120 , as well as an output RF signal  121 . The feedback currents  119  and  120  are coupled to the output windings of feedback transformers  107  and  108 , respectively, thereby forming a pair of lossless feedback amplifiers which serve to establish the conversion gain and improve the IM performance of the mixer. 
     Those familiar with the art will readily understand that the improved NF performance of the lossless feedback double-balanced active mixer is a result of the lack of additional noise sources in the embedding topology. This active mixer offers considerable advantages over the more traditional Gilbert Cell active mixer, especially in terms of signal-handling and performance variations over temperature due to the temperature dependency of the emitter resistance r e  of the driver transistors, and the tradeoffs that are encountered in receiver and transmitter system design. It further provides an improvement in NF over the Gilbert Cell mixer and the series-shunt feedback mixer. 
     It is the purpose of this invention to further advance the art of feedback mixers by addressing the sources of noise present in the lossless feedback double-balanced active mixer, and to therefore provide an active mixer of substantially improved NF performance, while at the same time retaining the desireable power consumption, IM performance, and overall sense of simplicity and cost effectiveness of the lossless feedback double-balanced active mixer. 
     SUMMARY OF THE INVENTION 
     A lossless feedback double-balanced active mixer circuit with improved intermodulation (IM) and noise figure (NF) performance is described which includes a pair of lossless feedback balanced active mixer circuits, each of which includes a differential pair of switching transistors which divide a controlled current into two paths at a rate determined by an input local oscillator (LO). A hybrid transformer in each lossless feedback balanced mixer, consisting of a centre-tapped primary winding and a secondary winding, combines the two currents to provide a recombined amplified IF signal and an output radio frequency (RF) signal. The controlled current is provided by an input intermediate frequency (IF) signal and its relation with the lossless feedback mixer input impedance. Each lossless feedback active mixer circuit further includes a feedback transformer, comprised of an input winding and a tapped output winding, which compares the input IF signal with the recombined amplified IF signal from the hybrid transformers and applies the difference as a correction to the input current, thereby completing a lossless feedback amplifier circuit which improves the IM performance of the mixer circuit. Since the feedback transformer is essentially lossless, it introduces no significant sources of noise to the active mixer circuit, and therefore the NF of the of the lossless feedback active mixer circuit remains unimpaired beyond the NF of the transistors themselves. The NF of the lossless feedback double-balanced active mixer is further improved by minimizing the number of active devices. The connection of the secondary windings of the hybrid transformers of the lossless feedback active mixer circuits effectively cancels the output LO and IF signals and provides and output RF signal. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The invention is described in the schematics of FIGS. 1 to  3 , in which: 
     FIG. 1 schematically illustrates the existing prior art, commonly referred to as a lossless feedback double-balanced active mixer; 
     FIG. 2 schematically illustrates a hybrid transformer; and 
     FIG. 3 schematically illustrates a low-noise lossless feedback double-balanced active mixer in accordance with the present invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     Designers of radio communication receivers are always concerned with elements of system performance which includes, but is not limited to, intermodulation distortion (IM), noise figure (NF), and power consumption. Historically, the IM performance of communications receivers is improved by methods that invariably require additional power consumption. Amplification stages with feedback methods are widely used as they offer far better IM performance while consuming less power than those not employing feedback. The NF of communications receivers is determined by the NF performance of the first stages of the receiver, which usually have sufficiently low NF and suitable signal gain to overcome the IM and NF performance of the first mixer stage, which is traditionally the primary source of distortion and noise. This invention now presents a mixer circuit which achieves a markedly improved IM and NF performance without excessive power consumption by applying a feedback method widely used in amplifier design which introduces no significant noise sources in addition to those of the active devices themselves. In addition, this invention removes a significant source of noise from prior embodiments. 
     A typical lossless feedback double-balanced active mixer circuit  100  is shown in FIG.  1 . Here, transistor  103  and transformer  107  form a lossless feedback amplifier on the left side, while transistor  106  and transformer  108  form a lossless feedback amplifier on the right side. Transistors  101  and  102  form a chopper for the left side and transistors  104  and  105  form a chopper for the right side. Hybrid transformer  111  combines currents  115  and  116  from transistors  101  and  102 , the sum of which appears at a centre tap while the difference appears at a secondary winding. A similar description can be made for the second hybrid transformer  112  on the right side. A differential input Intermediate Frequency (IF) signal connected to the input windings of transformers  107  and  108  generates a differential pair of input currents  113  and  114 :                I   113     =       I   Q     +       A   ×   Cos                   ω   S        t       R   in                 (   1   )                 I   114     =       I   Q     -       A   ×   Cos                   ω   S        t       R   in                 (   2   )                                
     where ω S  is the frequency and A is the amplitude of the input IF signal, I Q  is the quiescent bias current, and R in  is the input resistance which is defined as:                R   in     =         M   +   N   +   1       M   2       ×     R   11               (   3   )                                
     where M and N are the turns ratios of the output windings of transformers  107  and  108 . These input currents are conducted to the emitters of a pair of driver transistors  103  and  106 , respectively, which in turn conduct the current to a first differential pair of switching transistors  101  and  102  and a second differential pair of switching transistors  104  and  105 . A Local Oscillator (LO) signal applied differentially across the base terminals of the differential switching transistor pairs results in two differential pairs of output currents:                I   115     =         I   113     ×       1   -     Cos                   ω   L        t       2       =           I   Q     ×     (     1   -     Cos                   ω   L        t       )       2     +       A   ×     [       Cos                   ω   S        t     -           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t       2       ]         2   ×     R   in                     (   4   )                 I   116     =         I   113     ×       1   +     Cos                   ω   L        t       2       =           I   Q     ×     (     1   +     Cos                   ω   L        t       )       2     +       A   ×     [       Cos                   ω   S        t     +           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t       2       ]         2   ×     R   in                     (   5   )                 I   117     =         I   114     ×       1   +     Cos                   ω   L        t       2       =           I   Q     ×     (     1   +     Cos                   ω   L        t       )       2     -       A   ×     [       Cos                   ω   S        t     +           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t       2       ]         2   ×     R   in                     (   6   )                 I   118     =         I   114     ×       1   -     Cos                   ω   L        t       2       =           I   Q     ×     (     1   -     Cos                   ω   L        t       )       2     -       A   ×     [       Cos                   ω   S        t     -           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t       2       ]         2   ×     R   in                     (   7   )                                
     where ω S  is the frequency of the input LO signal. 
     Referring now to FIG. 2, a circuit  200  is used as an aid in describing the impedances, voltages, and currents of the four ports of a hybrid transformer  201 , which are: 
     
       
           R   204 =K 2   ×R   202   (8) 
       
     
     
       
           R   203   =R   205 =2 ×R   204   (9) 
       
     
     
       
           I   206 =K×( I   209   −I   207 )  (10) 
       
     
     
       
           I   208   =I   209   +I   207   (11) 
       
     
     
       
         
           
             
               
                 
                   
                     V 
                     206 
                   
                   = 
                   
                     
                       
                         V 
                         209 
                       
                       - 
                       
                         V 
                         207 
                       
                     
                     
                       2 
                       × 
                       K 
                     
                   
                 
               
               
                 
                   ( 
                   12 
                   ) 
                 
               
             
             
               
                 
                   
                     V 
                     208 
                   
                   = 
                   
                     
                       
                         V 
                         209 
                       
                       + 
                       
                         V 
                         207 
                       
                     
                     2 
                   
                 
               
               
                 
                   ( 
                   13 
                   ) 
                 
               
             
           
         
                 
         
             
         
      
     
     If both hybrid transformers  111  and  112  have turns ratios of 1:1:1 (K=1), then the currents at the center taps of the hybrid transformers  111  and  112  are, respectively:                I   119     =         I   115     +     I   116       =       I   Q     +       A   ×   Cos                   ω   S        t       R   in                   (   14   )                 I   120     =         I   117     +     I   118       =       I   Q     -       A   ×   Cos                   ω   S        t       R   in                   (   15   )                                
     and the output signal current conducted to the load resistance R L  is:                i   121     =         K   ×     (       I   115     -     I   116       )       -     K   ×     (       I   117     -     I   118       )         =     2   ×   A   ×     K   2     ×           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t         R   in                   (   16   )                                
     which makes the output signal voltage equal to:                v   121     =     2   ×   A   ×     K   2     ×     R   L     ×           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t         R   in                 (   17   )                                
     If the gain and output noise power of the driver transistors is constant across all frequencies, the square-wave switching process of the differential switching transistors would increase the input-referred noise contribution from the driver stage by a factor of (π/2) 2 , or 3.9 dB. This is a result of the square-wave LO mixing noise at various IF (of RF) frequencies up to the RF (or down to the IF), and on a linear scale the overall noise power of the mixer would be: 
     
       
           NF=N   D ×({fraction (π/2)}) 2   +N   SW   (18) 
       
     
     where N SW  is the noise contribution from the first differential switching pair  101  and  102  and the second differential switching pair  104  and  105 , and N D  is the input-referred noise of the driver transistors  103  and  106 , which consists of base shot noise (N b ), collector shot noise (N c ), and thermal noise (N t ): 
     
       
           N   D =1 +N   c   +N   b   +N   t   (19) 
       
     
     By examination of EQ. 18, if the sources of noise attributed to the driver transistors  103  and  106 , as described in EQ. 19, were to be reduced or eliminated, the NF of the lossless feedback double-balanced active mixer would be reduced to the NF of the differential pair switching transistors  101 ,  102 ,  104 , and  105 . 
     Here it should be noted that differential switching transistors  101  and  102 , driver transistor  103 , hybrid transformer  111 , and feedback transformer  107  form a first lossless feedback balanced active mixer circuit. Switching transistors  104  and  105 , driver transistor  106 , hybrid transformer  112 , and feedback transformer  108  forma a second lossless feedback balanced active mixer circuit. The lossless feedback double-balanced active mixer circuit  100  is formed by the combination of the first lossless feedback balanced active mixer circuit with the second lossless feedback balanced active mixer circuit. 
     Referring now to FIG. 3, a lossless feedback double-balanced active mixer circuit  300  in accordance with the present invention is illustrated. Mixer circuit  300  includes a first pair of switching transistors  301  and  302  and a second pair of switching transistors  303  and  304 . The emitters of switching transistors  301  and  302  are connected in common to one end of an input winding of a first lossless feedback transformer  305 , the opposite end of which is connected to receive one of a complementary pair of IF signals thereon. The emitters of switching transistors  303  and  304  are connected in common to one end of an input winding of a second lossless feedback transformer  306 , the opposite end of which is connected to receive the other of the complementary pair of IF signals thereon. The bases of switching transistors  301  and  304  are connected together to receive one of a complementary pair of local oscillator signals thereon. The bases of switching transistors  302  and  303  are connected together to receive the other of the complementary pair of local oscillator signals thereon. 
     The collectors of switching transistors  301  and  302  are connected to opposite sides of a primary winding of a first hybrid transformer  309 . A centre tap of the primary winding is connected to one end of an output winding of lossless feedback transformer  305 . The opposite end of the output winding is connected to a voltage source V cc  and a tap of the output winding is connected through a load resistance  307  (illustrated as a fixed resistance R 31  for convenience) to voltage source V cc . The collectors of switching transistors  303  and  304  are connected to opposite sides of a primary winding of a hybrid transformer  310 . A centre tap of the primary winding is connected to one end of an output winding of lossless feedback transformer  306 . The opposite end of the output winding is connected to a voltage source V cc  and a tap of the output winding is connected through a load resistance  308  (illustrated as a fixed resistance R 31  for convenience) to voltage source V cc . An RF output terminal  319  is connected through a secondary winding of hybrid transformer  309  to ground, through a secondary winding of hybrid transformer  310  to ground, and through a load resistance  320  (illustrated as a fixed resistance R L  for convenience) to ground. 
     Thus, a first low-noise lossless feedback balanced active mixer circuit includes switching transistors  301  and  302 , hybrid transformer  309 , and feedback transformer  305  and a second low-noise lossless feedback balanced active mixer circuit includes switching transistors  303  and  304 , hybrid transformer  310 , and feedback transformer  306 . Further, the low-noise lossless feedback double-balanced active mixer circuit  300  is formed by the combination of the first low-noise lossless feedback balanced active mixer circuit and the second low-noise lossless feedback balanced active mixer circuit. 
     The input impedance of mixer circuit  300 , as seen at either of the IF input ports, is determined from the value of the resistors  307  and  308 , as well as the turns ratios M and N of the lossless feedback transformers  305  and  306 :                R   in     =         M   +   N   +   1       M   2       ×     R   31               (   20   )                                
     It is necessary that the impedance of the centre tap of hybrid transformers  309  and  310  be matched to the collector load impedance of the lossless feedback transformers  305  and  306 , respectively: 
       R   317 =( M+N )× R   31 =2 ×K   2   ×R   L   (21) 
     
       
         
           
             
               
                 
                   
                     R 
                     31 
                   
                   = 
                   
                     
                       2 
                       × 
                       
                         K 
                         2 
                       
                       × 
                       
                         R 
                         L 
                       
                     
                     
                       M 
                       + 
                       N 
                     
                   
                 
               
               
                 
                   ( 
                   22 
                   ) 
                 
               
             
           
         
                 
         
             
         
      
     
     which forces the IF input impedance of both sides of the double-balanced lossless feedback active mixer circuit to be:                R   in     =       2   ×     K   2     ×     R   L     ×     (     M   +   N   +   1     )           M   2     ×     (     M   +   N     )                 (   23   )                                
     These conditions being satisfied, the input currents to the common emitter of the first differential switching transistor pair  301  and  302  and the second differential switching transistor pair  303  and  304  are, respectively:                I   311     =       I   Q     +       A   ×   Cos                   ω   S        t       R   in                 (   24   )                 I   312     =       I   Q     -       A   ×   Cos                   ω   S        t       R   in                 (   25   )                                
     where I Q  is the quiescent bias current, and A and ω S  are the amplitude and frequency, respectively, of the input IF (or RF) signal voltage. The current at the collectors of switching transistors  301 ,  302 ,  303 , and  304  are, respectively:                I   313     =         I   311     ×       1   -     Cos                   ω   L        t       2       =           I   Q     ×     (     1   -     Cos                   ω   L        t       )       2     +       A   ×     [       Cos                   ω   S        t     -           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t       2       ]         2   ×     R   in                     (   26   )                 I   314     =         I   311     ×       1   +     Cos                   ω   L        t       2       =           I   Q     ×     (     1   +     Cos                   ω   L        t       )       2     +       A   ×     [       Cos                   ω   S        t     +           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t       2       ]         2   ×     R   in                     (   27   )                 I   315     =         I   312     ×       1   +     Cos                   ω   L        t       2       =           I   Q     ×     (     1   +     Cos                   ω   L        t       )       2     -       A   ×     [       Cos                   ω   S        t     +           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t       2       ]         2   ×     R   in                     (   28   )                 I   316     =         I   312     ×       1   -     Cos                   ω   L        t       2       =           I   Q     ×     (     1   -     Cos                   ω   L        t       )       2     -       A   ×     [       Cos                   ω   S        t     -           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t       2       ]         2   ×     R   in                     (   29   )                                
     The currents at the centre taps of hybrid transformers  309  and  310  are, respectively:                I   317     =         I   313     +     I   314       =       I   Q     +       A   ×   Cos                   ω   S        t       R   in                   (   30   )                 I   318     =         I   315     +     I   316       =       I   Q     -       A   ×   Cos                   ω   S        t       R   in                   (   31   )                                
     and the current is:                i   319     =         K   ×     (       I   313     -     I   314       )       -     K   ×     (       I   315     -     I   316       )         =     2   ×   A   ×     K   2     ×           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t         R   in                   (   32   )                                
     which makes the output voltage equal to:                v   319     =     2   ×   A   ×     K   2     ×     R   L     ×           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t         R   in                 (   33   )                                
     which is identical to EQ. 16 and EQ. 17, respectively, showing that the low-noise lossless feedback double-balanced active mixer circuit  300  has the same conversion gain properties as the lossless double-balanced active mixer circuit  100  while the sources of noise have been substantially reduced. 
     Although detailed embodiments of the invention have been described, it should be appreciated that numerous modifications, variations, and adaptations may be made without departing from the scope of the invention as described in the claims. For example, those familiar with the art will recognize that the bipolar transistors shown in the embodiments may be alternatively replaced with field effect transistors. Also, the single-transformer lossless feedback topology shown in the embodiments may be alternatively replaced with other forms of lossless feedback that are known to the art. 
     Further, while the terminals of the bipolar transistors described in the various embodiments are referred to as the emitter, base, and collector, it will be understood that these terminals will be the source, gate, and drain when the transistors utilized are field effect transistors or other similar types and may be referred to as input, control, and output terminals, respectively, however the titles of the various components and terminals are only intended to enhance the understanding of the disclosure and are not intended to in any way limit the type of component utilized. In addition, it should be understood that the terms “lossless feedback transformer” and “hybrid transformer” used throughout this disclosure refer to general types of transformers and should not be limited in any way to specific types of transformers.