Abstract:
The invention relates to position sensors of the linearly variable induction difference type. When cost constraints prevent the use of transformers with guaranteed phase-shift tolerance to achieve an accuracy objective, it is advantageous to provide an independent demodulation of the signals of the two windings. The error signal thus has a lower dependence on the phase shift and the accuracy is typically enhanced by a factor greater than an order of magnitude.

Description:
FIELD OF THE INVENTION 
       [0001]    The present invention applies to the devices and methods for processing signals at the output of position sensors, notably of the linearly variable induction difference type. These sensors are generally designated by their English name Linear Variable Differential Transformer, or LVDT. 
       BACKGROUND OF THE INVENTION 
       [0002]    The sensors of this type normally consist of a transformer comprising a primary circuit to which is supplied an alternating current and two secondary circuits in which a ferromagnetic part in linear motion generates signals, the demodulation of which will enable the measurement of the displacement of the moving part to be acquired. These sensors and their conditioning electronics can have numerous applications: monitoring works of art, monitoring the production of mechanical parts, measuring the level of a liquid in tanks, monitoring the position of vehicle controls, for example a motor vehicle, a ship or an aircraft. The processing of the signal can differ according to the accuracy and the reliability sought for a given application. 
         [0003]    One of the main problems is the phase shift that appears between the signals of the two secondaries which affects the accuracy of the measurement when a conventional synchronous demodulation is applied. One of the known responses is to use transformers with guaranteed phase shift tolerance. However this adds significantly to the cost of the LVDTs, which can be prohibitive in the case of acquisition subsystems with several tens of LVDTs which are commonly used in aeronautics. 
       SUMMARY OF THE INVENTION 
       [0004]    The aim of the present invention is to resolve this problem by considerably reducing the inaccuracies resulting from the phase shifting of the secondary windings, and therefore without the use of components with guaranteed tolerance. Although it applies to the processing of signals from any type of LVDT, embodiments of the present invention may be used to monitor aircraft flight controls, for which the prior art requires costly circuits to meet specification requirements. 
         [0005]    To this end, the present invention proposes a device for decoding signals at the output of two secondary coils in the axis of which is displaced a ferromagnetic part excited by a primary coil comprising a module for converting said signals from analogue to digital, a module for multiplying the digitized signals by chosen factors, a module for loop-calculating the error on the position of the magnetic part from signals at the output of the multiplication module, wherein said error calculation module comprises two synchronous demodulation channels each applied to one of the error signals specific to one of the secondary coils. 
         [0006]    It also proposes a method of using said device. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0007]    The invention will be better understood and its different characteristics and advantages will become apparent from the following description of a number of exemplary embodiments, and its appended figures in which: 
           [0008]      FIG. 1  is a general diagram of an LVDT; 
           [0009]      FIG. 2  represents the signal at the output of a device for processing the signals at the output of an LVDT; 
           [0010]      FIG. 3  represents the generation of the error signals on the two channels of a device according to one embodiment of the invention; 
           [0011]      FIG. 4  represents the schematic diagram of a dual integration loop according to one embodiment of the invention; 
           [0012]      FIG. 5  represents the schematic diagram of the generation of the error signals on the two demodulation channels according to the invention. 
       
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       [0013]    In the three  FIGS. 1.1 ,  1 . 2  and  1 . 3 , a linear variable differential transformer position sensor  100  is represented in three different positions. It comprises a generator  110  of alternating current feeding a primary winding  120  called excitation signal. A signal is created in the two secondary windings  130  and  140  by the displacement of the ferromagnetic core  150  which is joined to the part for which the displacement is to be measured. In this embodiment, the two windings are mounted in series but in opposition so that it is the difference of the currents in the secondary windings that is measured. In  FIG. 1.1 , the core is at the maximum of its left travel and the current at the output of the pair of secondaries  130 ,  140  is equal to the difference of the current at the terminals of the left secondary winding  130  and of the current at the terminals of the right secondary winding  140 . In  FIG. 1.2 , the core is at rest, in a neutral position and the output current from the pair of secondaries  130 ,  140  is zero. In  FIG. 1.3 , the core is at the maximum of its right travel and the current at the output of the pair of secondaries  130 ,  140  is equal to the difference of the current at the terminals of the right secondary winding  140  and of the current at the terminals of the left secondary winding  130 . There is a wide variety of LVDTs for measuring travel from a few micrometres to a few tens of centimetres. The travel of the core is bounded so that the variations of the current are proportional to its displacements. 
         [0014]      FIG. 2  shows the characteristic quantities of the operation of the sensor.  FIG. 2.1  represents the variation of the amplitude of the differential analogue signal induced in the secondary circuit by the excitation of the primary winding according to the travel of the core.  FIG. 2.2  represents the phase shift of the signal from the secondary circuit relative to the excitation signal, once again according to the travel of the core. The accuracy of the acquisition of the position data by an LVDT therefore depends notably on the excitation frequency of the signal in the primary circuit—which must be chosen to minimize the noise in the measurement subsystem—and the quality of the demodulation of the signal at the output of the sensor. 
         [0015]    An accurate synchronous demodulation can be achieved by using a type II locked loop, that is a dual-integration locked loop, the general operating principle of which is explained hereinafter in the description with the following notations: 
         [0000]    X in : travel of the core of the LVDT at the loop input;
 
X out : travel of the core of the LVDT at the loop output;
 
V 1 : voltage of the signal at the output of the secondary winding  130 ;
 
V 2 : voltage of the signal at the output of the secondary winding  140 ;
 
E 0 : peak amplitude of the signals on the secondaries;
 
X 0 : maximum value of the travel X in  of the core;
 
f: excitation frequency of the primary winding (also use ω=2πf);
 
Err: travel measurement error;
 
φ 0 : phase shift between V 1  and V 2 .
 
         [0016]    In the theoretical case of an absence of phase shift, the signals V 1  and V 2  are expressed: 
         [0000]        V 1=(½)(1 +X   in   /X   0 )· E   0 ·sin(ω t ) 
         [0000]        V 2=(½)(1− X   in   /X   0 )· E   0 ·sin(ω t ) 
         [0017]    V 1  and V 2  are both digitized by analogue/digital converters (ADC). To allow for the error signal to be calculated easily, as indicated in  FIG. 3 , they are respectively multiplied by: 
         [0000]      λ 1 =1 −X   out   /X   0  and 
         [0000]      λ 2 =1 +X   out   /X   0    
         [0018]    so as to create two error signals Err 1  and Err 2  with respective values λ 1 V 1  and λ 2 V 2 . In the prior art, the error signal is created by obtaining the difference between Err 1  and Err 2 . This error signal is then demodulated synchronously by using the excitation signal as a reference. 
         [0000]        Err =(½) E   0 ·sin(ω t )·((1 +X   in   /X   0 )·(1− X   out   /X   0 )−(1 −X   in   /X   0 )·(1 +X   out   /X   0 )) 
         [0000]      Or  Err=E   0 ·sin(ω t )·( X   in   /X   0   −X   out   X   0 ) 
         [0019]    A demodulation loop is represented in  FIG. 4  where the parameters and expressions have the following meanings and, by way of illustration, the following values: 
         [0000]                                                Input ADC gain   k 1  = V IN /V REF                 V REF : ADC reference voltage           Gain on error   k 2  = 18 × 10 6  × 2π           Filter zero   a = 4095/4096           Filter pole   b = 4085/4096           Integrator gain   c = 1/4096000           Integrator transfer function   I(z) = c/(1 − z −1 )           Filter transfer function   C(z) = (1 − az −1 )/(1 − bz −1 )           Open loop transfer function   G(z) = k 1  · k 2  · I 2 (z) · C(z)           Closed loop transfer function   H(z) = G(z)/(1 + G(z))                        
The loop cancels the error signal with the accuracy of the converter. It is designed to follow without error an input position which changes at constant speed.
 
         [0020]    If there is a phase shift φ 0  of V 2  relative to V 1 , V 2  is rewritten: 
         [0000]        V 2=(½)(1 −X   in   /X   0 )· E   0 ·sin(ω t+φ   0 ) 
         [0000]    And the expression of the error is as follows: 
         [0000]        Err =(½)[ E   0 ·sin(ω t )·((1 +X   in   /X   0 )·(1 −X   out   /X   0 )) 
         [0000]      −E 0 ·sin(ωt+φ 0 )·((1−X in /X 0 )·(1+X out /X 0 ))] or 
         [0000]        Err=E   0 ·[sin(ω t )·[(1−cos φ 0 )·(1 −X   in   ·X   out   /X   0   2 )+(1+cos φ 0 )·( X   in   /X   0   −X   out   /X   0 )]−cos(ω t )sin φ 0 (1 −X   in   /X   0 )·(1 +X   out   /X   0 )] 
         [0021]    After demodulation, the term which is a function of cos(ωt) is eliminated because it is in quadrature and we have: 
         [0000]        Err demod= KE   0 ·[(1−cos φ 0 )·(1 −X   in   ·X   out   /X   0   2 )+(1+cos φ 0 )·( X   in   /X   0   −X   out   /X   0 )] 
         [0000]    expression in which K is a given factor for a chosen setting of the loop.
 
The calculation shows that this error is cancelled for X out  equal to X in +δX with δX/Xo equal to:
 
         [0000]      δ X/Xo =(1−cos φ 0 )·(1 −X   in   2   /X   0   2 )/[(1+cos φ 0 )+ X   in   /X   0 (1−cos φ 0 )] 
         [0000]    The error is maximum for X in  equal to 0.
 
In this case δX/Xo=(1−cos φ 0 )/(1+cos φ 0 )
 
         [0022]    For φ 0  equal to 10°, the error is 0.8% which is prohibitive in view of the required accuracies. One simple but costly solution to this accuracy inadequacy is to use components with phase shifts guaranteed to be less than 3°. The invention makes it possible to use components with more relaxed phase shift tolerances. The principle of the invention is to limit the weighting of the phase shift in the calculation of the error by calculating the latter only after independent demodulation of the two channels. 
         [0023]    As illustrated in  FIG. 3 , two error signals Err 1  and Err 2  are therefore extracted after multiplying V 1  and V 2  respectively by λ 1  and λ 2 . 
         [0024]    As illustrated in  FIG. 5 , these two error signals Err 1  and Err 2  are then demodulated by two independent channels, the dual integration loop being of the same type as that illustrated in  FIG. 4 , the operation of which has already been described hereinabove. The overall error is then calculated by the difference of the two channels, as illustrated on the right of  FIG. 5 . This independent calculation of the errors is possible in the case of the LVDT because, by definition, |X in | and |X out | are always less than |X 0 |. Therefore λ 1  and λ 2  are always positive, Err 1  has the same sign as sin(ωt) and Err 2  the same sign as sin(ωt+φ 0 ). The demodulation therefore consists in multiplying Err 1  by +1 when Err 1  is positive and by −1 when it is negative. Similarly, Err 2  is multiplied by +1 when Err 2  is positive and by −1 when it is negative. The difference of the two rectified errors is then obtained and integrated. 
         [0025]    In this way, the errors due to the phase shifts between primary and secondary and between secondaries imparted by the sensor are in principle cancelled. In effect, the two full-wave rectifications eliminate on the one hand the term which is a function of sin(ωt) of Err 1  and on the other hand the term which is a function of sin(ωt+φ 0 ) of Err 2 . The expression of the total demodulated error therefore takes the form: 
         [0000]        Err demod= K ′(( X   in   /X   0   −X   out   /X   0 ) 
         [0000]    When the loop converges (X out =X in ), the error is therefore cancelled. 
         [0026]    Simulations have been carried out for different phase-shift values with a simple demodulation after error calculation (Case 1) and with dual demodulation of the errors (Case 2). The residual errors obtained in these simulations are given in the table below and fully confirm the advantage provided by the invention since, in the intermediate case, the gain in accuracy is by a factor of 18. 
         [0000]                                    Phase shift   Accuracy (Case 1)   Accuracy (Case 2)                    3°   0.11%    0.02%       10°   0.9%   0.05%       20°   3.4%    0.1%                    
The duplication of the demodulation subsystem only very slightly increases the resources needed in a programmable circuit or an ASIC for a very significant benefit on performance in the presence of significant phase shift between the two inputs.