Abstract:
The present invention includes: an input voltage detector detecting an input voltage of a parallel converter and outputting an input voltage signal; an output voltage detector detecting an output voltage of the parallel converter; and a controller. The controller includes an error amplifier comparing the output voltage signal and a reference voltage and outputting an error amplification signal; an arithmetic operator generating an ON time signal and an OFF time signal based on the input and output voltage signals and the error amplification signal; a phase signal generator generating plural phase signals having different phases based on the ON and OFF time signals and the error amplification signal; a pulse generator generating plural pulse-train signals synchronized with the respective phase signals based on the ON time signal, the error amplification signal and the phase signals; and a driver driving the switching units in accordance with the pulse-train signals.

Description:
TECHNICAL FIELD 
     The present invention relates to an interleaved converter, and particularly, to a technique of a controller for the interleaved converter. 
     BACKGROUND ART 
     An interleaved converter is disclosed in, for example, Japanese Patent Application Publication Nos. Sho 62-58871 and Sho 63-186555 as well as Japanese Patent No. 3570113. The interleaved converter is an electric power conversion device in which multiple converters are connected in parallel and are phase-shifted from one another to reduce current ripples of the current inputted to the converter and the current to be outputted from the converter. In addition, a phase controller for an interleaved converter is disclosed in Japanese Patent Nos. 3570113 and 3480201. 
       FIG. 1  is a circuit configuration diagram showing a conventional interleaved converter using two circuits of boost converters. 
     In  FIG. 1 , a first series circuit, which includes a boost reactor L 1 , a switching element Q 1  made of a MOSFET, and a switching current detector CT 1 , is connected to the terminals of an input power source Vin formed of a DC power source. The anode of a rectifier D 1  is connected to a connection point of the reactor L 1  and the switching element Q 1  while the cathode of the rectifier D 1  is grounded through a smoothing capacitor Co. 
     A second series circuit, which includes a boost reactor L 2 , a switching element Q 2  made of a MOSFET, and a switching current detector CT 2 , is connected to the terminals of the input power source Vin. The anode of a rectifier D 2  is connected to a connection point of the boost reactor L 2  and the switching element Q 2  while the cathode of the rectifier D 2  is grounded through the smoothing capacitor Co. 
     A voltage detector  20  is configured to receive an output voltage Vo outputted from the terminals of the smoothing capacitor Co, and output an output voltage signal VFB. A first control circuit  21  is configured to generate an output signal Vdr 1  based on an output from the switching current detector CT 1  and the output voltage signal VFB, and thereby perform on/off control of the gate of the switching element Q 1  by using the output signal Vdr 1 . A charge/discharge device  23  is configured to receive the output signal Vdr 1  from the first control circuit  21 . A phase control capacitor C 21  is connected to one of output terminals of the charge/discharge device  23  and a phase control capacitor C 22  is connected to the other one of the output terminals of the charge/discharge device  23 . 
     A second control circuit  22  is configured to generate an output signal Vdr 2  based on an output Vi 2  from the switching current detector CT 2 , the output voltage signal VFB, as well as outputs Vc 1  and Vc 2  of the phase control capacitors C 21  and C 22 , and thereby perform on/off control of the gate of the switching element Q 2  by using the output signal Vdr 2 . 
     The boost reactor L 1 , the switching element Q 1 , the switching current detector CT 1 , the rectifier D 1 , and the first control circuit  21  form a first converter. The boost reactor L 2 , the switching element Q 2 , the switching current detector CT 2 , the rectifier D 2 , and the second control circuit  22  form a second converter. The first converter and the second converter are connected to each other at the respective input terminals as well as at the respective output terminals, thereby forming a boost interleaved converter. 
     The boost converter is configured to output an output voltage Vo that is higher than an input voltage Vin in accordance with the ON/OFF operations of the switching elements Q 1  and Q 2 . When the switching element Q 1  (or Q 2 ) is ON, current flows from Vin, through L 1  (or L 2 ) and then Q 1  (or Q 2 ), to Vin, so that an energy of the magnetic flux is accumulated in the boost reactor L 1  (or L 2 ). When the switching element Q 1  (or Q 2 ) is OFF, current flows from Vin through L 1  (or L 2 ), D 1  (or D 2 ), and Co, to Vin, so that the energy of the magnetic flux in the boost reactor L 1  (or L 2 ), which is accumulated during the ON time of the switching element Q 1  (or Q 2 ), is discharged. This operation is expressed by the following expression: 
     
       
         
           
             
               
                 
                   
                     Δ 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     IL 
                   
                   = 
                   
                     
                        
                       
                         
                           Vin 
                           L 
                         
                         · 
                         Ton 
                       
                        
                     
                     ≤ 
                     
                       
                          
                         
                           
                             
                               Vo 
                               + 
                               VF 
                               - 
                               Vin 
                             
                             L 
                           
                           · 
                           Toff 
                         
                          
                       
                       . 
                     
                   
                 
               
               
                 
                   Λ 
                   ⁡ 
                   
                     ( 
                     1 
                     ) 
                   
                 
               
             
           
         
       
     
     In the expression (1), ΔIL represents an amount of change in current flowing through the boost reactor L 1  (or L 2 ), Vin the voltage across the input power source Vin, Vo the voltage across the smoothing capacitor Co, VF the forward drop voltage of the rectifier D 1  (or D 2 ), L the value of the inductance of the boost reactor L 1  (or L 2 ), Ton the ON time of the switching element Q 1  (or Q 2 ), and Toff the OFF time of the switching element Q 1  (or Q 2 ). The minimum value of the OFF time Toff of the switching element Q 1  (or Q 2 ) is obtained from the expression (2) using the input voltage Vin, the output voltage Vo, and the ON time Ton: 
     
       
         
           
             
               
                 
                   Toff 
                   ≥ 
                   
                     
                       Vin 
                       
                         Vo 
                         + 
                         VF 
                         - 
                         Vin 
                       
                     
                     · 
                     
                       Ton 
                       . 
                     
                   
                 
               
               
                 
                   Λ 
                   ⁡ 
                   
                     ( 
                     2 
                     ) 
                   
                 
               
             
           
         
       
     
       FIG. 2  is a chart showing the operation waveforms of components of the conventional interleaved converter. In  FIG. 2 , Vdr 1  represents the drive signal of the switching element Q 1 , Vc 21  and Vc 22  the voltages across the phase control capacitors C 21  and C 22 , respectively, Vdr 2  the drive signal of the switching element Q 2 , Vi 1  the output signal of the switching current detector CT 1 , Vi 2  the output signal of the switching current detector CT 2 , Ii 1  the current flowing through the reactor L 1 , Ii 2  the current flowing through the reactor L 2 , Ii the input current of the interleaved converter, Id 1  the current flowing through the rectifier D 1 , Id 2  the current flowing through the rectifier D 2 , and Io the output current of the interleaved converter. 
     The first converter outputs the drive signal Vdr 1  for driving the switching element Q 1 , based on the output signal VFB of the voltage detector  20  and the output signal Vi 1  of the switching current detector CT 1 , thereby converting the voltage from the input voltage Vin to the output voltage Vo. When the switching element Q 1  is turned ON, the input voltage Vin is applied to the boost reactor L 1 , so that an energy of the magnetic flux is accumulated in the boost reactor L 1 . When the switching element Q 1  is turned OFF, the energy of the magnetic flux accumulated in the boost reactor L 1  is charged to the smoothing capacitor Co through the rectifier D 1 . 
     In this manner, the first converter performs the power conversion through the path from the input power source Vin to the smoothing capacitor Co by use of the ON/OFF operations of the switching element Q 1 . In the same manner, the second converter performs the power conversion through the path from the input power source Vin to the smoothing capacitor Co by use of the ON/OFF operations of the switching element Q 2 . Operating the first converter and the second converter respectively with mutually different phases suppresses the ripple of the current flowing through the input power source Vin and the smoothing capacitor Co. The suppressing effect is proportional to the number of converters connected in parallel with a constant phase difference between each two of the converters. 
     In the conventional example shown in  FIG. 1 , the charge/discharge device  23  is provided to appropriately control the phase of each converter. The charge/discharge device  23  is configured to perform the charging and discharging of the phase control capacitors C 21  and C 22  in synchronization with the drive signal Vdr 1  of the switching element Q 1 . When the phase control capacitor C 21  is charged, the phase control capacitor C 22  is discharged. When the phase control capacitor C 21  is discharged, the phase control capacitor C 22  is charged. The voltages Vc 21  and Vc 22  respectively across the phase control capacitors C 21  and C 22  form triangular waves having phases mutually shifted by 180 degrees. From the comparison of signals of the two triangular waves having phases mutually shifted, it is found that the values of the voltages Vc 21  and Vc 22  cross each other at the middle of each cycle of the drive signal Vdr 1  of the switching element Q 1 . The second control circuit  22  compares the inputted voltages Vc 21  and Vc 22  respectively across the phase control capacitors C 21  and C 22  with each other, thereby detecting a time point at which the voltages Vc 21  and Vc 22  cross each other, and outputs the drive signal Vdr 2  to the switching element Q 2  at the detected time point. With such a configuration, the second converter is given a voltage with a phase which is different by 180 degrees from that of the first converter. 
     However, in the interleaved converter configured as described above, the phase difference between the first and second converters varies due to a difference in capacity between the phase control capacitors C 21  and C 22 . In addition, the interleaved converter requires phase control capacitors the number of which is equal to or larger than the number of converters connected in parallel. For this reason, an increase in the number of converters makes the circuit of the interleaved converter complicated. 
     An object of the present invention is thus to provide an inexpensive interleaved converter having a simplified circuit. 
     SUMMARY OF INVENTION 
     A first aspect of the present invention provides an interleaved converter including: a parallel converter including a plurality of converters connected in parallel, each of the plurality of converters including a reactor, a switching unit, and a rectifier; an input power source configured to supply power to the parallel converter; a smoothing capacitor configured to smooth an output of the parallel converter; an input voltage detector configured to detect an input voltage of the parallel converter, and thereby to output an input voltage signal; an output voltage detector configured to detect an output voltage of the parallel converter, and thereby to output an output voltage signal; and a controller configured to control the parallel converter. In the interleaved converter, the controller includes: an error amplifier configured to compare the output voltage signal with a reference voltage, and thereby to output an error amplification signal; an arithmetic operator configured to perform arithmetic processing based on the input voltage signal, the output voltage signal, and the error amplification signal, and thereby to generate an ON time signal and an OFF time signal; a phase signal generator configured to generate a plurality of phase signals having mutually different phases, based on the ON time signal, the OFF time signal, and the error amplification signal; a pulse generator configured to generate a plurality of pulse-train signals synchronized respectively with the plurality of phase signals, based on the ON time signal, the error amplification signal, and the plurality of phase signals; and a driver configured to drive the switching units in accordance with the plurality of pulse-train signals. 
     In addition, a second aspect of the present invention provides an interleaved converter including: a parallel converter including a plurality of converters connected in parallel, each of the plurality of converters including a reactor, a switching unit, and a rectifier; an input power source configured to supply power to the parallel converter; a smoothing capacitor configured to smooth an output of the parallel converter; an input voltage detector configured to detect an input voltage of the parallel converter, and thereby to output an input voltage signal; an output voltage detector configured to detect an output voltage of the parallel converter, and thereby to output an output voltage signal; and a controller configured to control the parallel converter. In the interleaved converter, the controller includes: an error amplifier configured to compare the output voltage signal with a reference voltage, and thereby to output an error amplification signal; an arithmetic operator configured to perform arithmetic processing based on the input voltage signal, the output voltage signal, and the error amplification signal, and thereby to generate an ON time signal and an OFF time signal; a phase signal generator configured to generate a plurality of phase signals having mutually different phases, based on the ON time signal and the OFF time signal; 
     a pulse generator configured to generate a plurality of pulse-train signals synchronized respectively with the plurality of phase signals, based on the ON time signal and the plurality of phase signals; and a driver configured to drive the switching units in accordance with the plurality of pulse-train signals. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG. 1  is a circuit configuration diagram showing an example of a conventional interleaved converter. 
         FIG. 2  is a chart showing the operation waveforms of components of the conventional interleaved converter. 
         FIG. 3  is a circuit configuration diagram showing an interleaved converter according to Embodiment 1 of the present invention. 
         FIG. 4  is a circuit configuration diagram showing an arithmetic operator provided in the interleaved converter according to Embodiment 1. 
         FIG. 5  is a circuit configuration diagram showing a multiplication and division circuit provided in the interleaved converter according to Embodiment 1. 
         FIGS. 6A and 6B  are graphs showing examples of waveforms indicating input/output characteristics of the arithmetic operator provided in the interleaved converter according to Embodiment 1. 
         FIG. 7  is a circuit configuration diagram showing a phase signal generator provided in the interleaved converter according to Embodiment 1. 
         FIG. 8  is a circuit configuration diagram showing a frequency divider circuit in the phase signal generator shown in  FIG. 7 . 
         FIG. 9  is a chart showing the operation waveforms of components of the phase signal generator shown in  FIG. 7 . 
         FIG. 10  is a circuit configuration diagram showing a pulse generator provided in the interleaved converter according to Embodiment 1. 
         FIG. 11  is a chart showing the operation waveforms of components of the pulse generator shown in  FIG. 10 . 
         FIG. 12  is a circuit configuration diagram showing an interleaved converter according to Embodiment 2 of the present invention. 
         FIG. 13  is a circuit configuration diagram showing an arithmetic operator provided in the interleaved converter according to Embodiment 2. 
         FIG. 14  is a circuit configuration diagram showing a phase signal generator provided in the interleaved converter according to Embodiment 2. 
         FIG. 15  is a circuit configuration diagram showing a frequency divider circuit in the phase signal generator shown in  FIG. 14 . 
         FIG. 16  is a chart showing the operation waveforms of components of the phase signal generator shown in  FIG. 14 . 
         FIG. 17  is a circuit configuration diagram showing a pulse generator provided in the interleaved converter according to Embodiment 2. 
     
    
    
     DESCRIPTION OF EMBODIMENTS 
     Hereinafter, an interleaved converter according to an embodiment of the present invention will be described in detail with reference to the accompanying drawings. This interleaved converter is configured to improve the power factor of power inputted from an input power source. 
     Embodiment 1 
       FIG. 3  is a circuit configuration diagram showing an interleaved converter according to Embodiment 1 of the present invention. A full-wave rectifier RC 1  is configured to receive an AC voltage from an AC power source Vac, full-wave rectify the received AC voltage, and thereby output a DC voltage Vin. A first series circuit, which includes a boost reactor L 1  and a switching element Q 1  made of a MOSFET, is connected to the output terminal of the full-wave rectifier RC 1 . The anode of a rectifier D 1  is connected to a connection point of the boost reactor L 1  and the switching element Q 1 , and the cathode of the rectifier D 1  is grounded through a smoothing capacitor Co. 
     A second series circuit, which includes a boost reactor L 2  and a switching element Q 2  made of a MOSFET, is connected to the output terminal of the full-wave rectifier RC 1 . The anode of a rectifier D 2  is connected to a connection point of the boost reactor L 2  and the switching element Q 2 , and the cathode of the rectifier D 2  is grounded through the smoothing capacitor Co. 
     A first voltage divider resistor including a resistor R 1  and a resistor R 2  is connected to the output terminal of the full-wave rectifier RC 1 . A second voltage divider resistor including a resistor R 3  and a resistor R 4  is connected to both terminals of the smoothing capacitor Co. 
     A control circuit  10  is configured to receive a midpoint voltage VIN of the first voltage divider resistor and a midpoint voltage VFB of the second voltage divider resistor, and to then output drive signals to the gates of the respective switching elements Q 1  and Q 2 . The control circuit  10  includes an error amplifier  11 , an arithmetic operator  12 , a phase signal generator  13 , a pulse generator  14 , and a drive circuit  15 . 
     The error amplifier  11  is configured to amplify an error between the midpoint voltage VFB of the second voltage divider resistor and a reference voltage Vref, and to thereby output an error amplification signal VCOMP. The arithmetic operator  12  is configured to receive the midpoint voltage VIN of the first voltage divider resistor, the midpoint voltage VFB of the second voltage divider resistor, and the error amplification signal VCOMP, to perform arithmetic operation on these voltages, and to thereby output an ON time signal Ion and an OFF time signal Ioff. The ON time signal Ion is a signal proportional to the ON time of the switching element Q 1  (or Q 2 ), and the OFF time signal Ioff is a signal proportional to the OFF time of the switching element Q 1  (or Q 2 ). 
     The phase signal generator  13  is configured to generate and output a phase signal Ph 1  and a phase signal Ph 2  having mutually different phases, based on the ON time signal Ion, the OFF time signal Ioff, and the error amplification signal VCOMP. The pulse generator  14  is configured to generate and output a pulse-train signal PWM 1  and a pulse-train signal PWM 2  having the same duty ratio and mutually different phases, based on the phase signal Ph 1 , the phase signal Ph 2 , the ON time signal Ion, and the error amplification signal VCOMP. The drive circuit  15  is configured to generate a first drive signal Vdr 1  for driving the switching element Q 1  and a second drive signal Vdr 2  for driving the switching element Q 2 , based on the pulse-train signals PWM 1  and PWM 2 , and to output the first drive signal Vdr 1  and the second drive signal Vdr 2  to the corresponding switching elements. 
     The boost reactor L 1 , the switching element Q 1 , and the rectifier D 1  form a first converter. The boost reactor L 2 , the switching element Q 2 , and the rectifier D 2  form a second converter. The first converter and the second converter are connected to each other at the respective input terminals as well as at the respective output terminals, thereby forming an interleaved converter. 
     The interleaved converter according to Embodiment 1 is configured to perform a power-factor improving operation to improve the power factor of an AC input current Iac inputted from an AC input power source Vac. The improving of the power factor requires for an input current to change in proportion to an input voltage. The state of the input current (IL·Vin) during the power-factor improvement being performed are expressed by the following expressions (3) and (4) obtained by modifying the expression (1): 
     
       
         
           
             
               
                 
                   
                     
                       IL 
                       · 
                       Vin 
                     
                     = 
                     
                       
                         
                           Vin 
                           L 
                         
                         · 
                         Ton 
                       
                       = 
                       
                         
                           
                             Vo 
                             + 
                             VF 
                             - 
                             Vin 
                           
                           L 
                         
                         · 
                         Toff 
                       
                     
                   
                   ; 
                 
               
               
                 
                   Λ 
                   ⁡ 
                   
                     ( 
                     3 
                     ) 
                   
                 
               
             
             
               
                 
                   IL 
                   = 
                   
                     
                       
                         1 
                         L 
                       
                       · 
                       Ton 
                     
                     = 
                     
                       
                         
                           Vo 
                           + 
                           VF 
                           - 
                           Vin 
                         
                         Vin 
                       
                       · 
                       
                         1 
                         L 
                       
                       · 
                       
                         Toff 
                         . 
                       
                     
                   
                 
               
               
                 
                   Λ 
                   ⁡ 
                   
                     ( 
                     4 
                     ) 
                   
                 
               
             
           
         
       
     
     The ON time of a switching element changes in accordance with power regardless of the phase of the input voltage. The OFF time of the switching element can be obtained using the input voltage, the output voltage, and the ON time of the switching element. 
     When the ON time signal Ton and the OFF time signal Toff of the switching element are generated in accordance with a constant current charge (or discharge) of a capacitor, the following expressions (5) and (6) are obtained: 
     
       
         
           
             
               
                 
                   
                     Ton 
                     = 
                     
                       
                         Cosc 
                         · 
                         VCOMP 
                       
                       Ion 
                     
                   
                   ; 
                 
               
               
                 
                   Λ 
                   ⁡ 
                   
                     ( 
                     5 
                     ) 
                   
                 
               
             
             
               
                 
                   Toff 
                   = 
                   
                     
                       
                         Cosc 
                         · 
                         VCOMP 
                       
                       Ioff 
                     
                     . 
                   
                 
               
               
                 
                   Λ 
                   ⁡ 
                   
                     ( 
                     6 
                     ) 
                   
                 
               
             
           
         
       
     
     In the expressions (5) and (6), Cosc represents an oscillator capacitor, VCOMP the error amplification signal, Ion the ON time signal, Ioff the OFF time signal. Since the ON time signal Ton of the switching element varies due to the error amplification signal VCOMP, the ON time signal Ion is a constant or a function of the error amplification signal VCOMP. When the OFF time signal Ioff is obtained by substituting the expressions (5) and (6) into the expression (2), the following expression (7) is obtained: 
     
       
         
           
             
               
                 
                   Ioff 
                   = 
                   
                     Ion 
                     ⁢ 
                     
                       
                         
                           Vo 
                           + 
                           VF 
                           - 
                           Vin 
                         
                         Vin 
                       
                       . 
                     
                   
                 
               
               
                 
                   Λ 
                   ⁡ 
                   
                     ( 
                     7 
                     ) 
                   
                 
               
             
           
         
       
     
     To put it differently, the OFF time signal Ioff is a value obtained by multiplying the ON time signal Ion by a value obtained by dividing a difference between the output voltage Vo and the input voltage Vin by the input voltage Vin. 
       FIG. 4  is a circuit configuration diagram showing the arithmetic operator  12  provided in the interleaved converter according to Embodiment 1 of the present invention. In  FIG. 4 , an operational amplifier AP 1  has a non-inverting input terminal connected to the error amplification signal VCOMP, an output terminal connected to the gate of a MOSFET Q 10 , and an inverting input terminal grounded through a resistor R 10  and connected to the source of the MOSFET Q 10 . A current signal Ivcomp proportional to the error amplification signal VCOMP is outputted from the drain of the MOSFET Q 10  to a multiplication and division circuit  122  through a first current mirror circuit  121 . The operational amplifier AP 1  and the MOSFET Q 10  form a first voltage/current converter circuit. 
     An operational amplifier AP 2  has a non-inverting input terminal connected to the midpoint voltage VIN of the first voltage divider resistor, an output terminal connected to the gate of a MOSFET Q 11 , and an inverting input terminal grounded through a resistor R 11  and connected to the source of the MOSFET Q 11 . A current signal Ivin proportional to the midpoint voltage VIN of the first voltage divider resistor is outputted from the drain of the MOSFET Q 11  to a multiplication and division circuit  123 . The operational amplifier AP 2  and the MOSFET Q 11  form a second voltage/current converter circuit. 
     An operational amplifier AP 3  has a non-inverting input terminal connected to the midpoint voltage VFB of the second voltage divider resistor, an output terminal connected to the gate of a MOSFET Q 12 , and an inverting input terminal grounded through a resistor R 12  and connected to the source of the MOSFET Q 12 . A current signal Ivfb proportional to the midpoint voltage VFB of the second voltage divider resistor is outputted from the drain of the MOSFET Q 12  to the multiplication and division circuit  123 . The operational amplifier AP 3  and the MOSFET Q 12  form a third voltage/current converter circuit. 
     In the first current mirror circuit  121 , the collector and the base of a transistor Q 13  as well as the base of a transistor Q 14  are connected to form an input terminal of the first current mirror circuit  121 . The emitters of the respective transistors Q 13  and Q 14  are connected to a power source Reg. The collector of the transistor Q 14  forms an output terminal of the first current mirror circuit  121 . The multiplication and division circuit  122  is configured to perform multiplication and division operations based on a constant current I 10  and the current signal Ivcomp inputted from the first current mirror circuit  121  and being proportional to the error amplification signal VCOMP, and to then output the ON time signal Ion as an output of the multiplication and division operations. 
     The multiplication and division circuit  123  is configured to perform multiplication and division operations based on the On time signal Ion from the multiplication and division circuit  122 , the current signal Ivin proportional to the midpoint voltage VIN of the first voltage divider resistor, and the current signal Ivfb proportional to the midpoint voltage VFB of the second voltage divider resistor, and to then output t+he OFF time signal Ioff as an output of the multiplication and division operations. 
       FIG. 5  is a circuit configuration diagram showing each of the multiplication and division circuits  122  and  123  provided in the interleaved converter according to Embodiment 1. In  FIG. 5 , the base and the collector of a transistor Q 30  are connected to the power source Reg, and the emitter of the transistor Q 30  is connected to an Ia input terminal and the base of a transistor Q 32 . The collector of the transistor Q 32  is connected to the power source Reg, and the emitter of the transistor Q 32  is connected to an Ib input terminal and the base of a transistor Q 34 . The emitter of the transistor Q 34  is connected to the emitter of a transistor Q 35  and one end of a current source Itail. 
     The collector of the transistor Q 34  is connected to the base of a transistor Q 33  and the emitter of a transistor Q 31 . The base of the transistor Q 35  is connected to an Ic input terminal and the emitter of the transistor Q 33 . The collector of the transistor Q 35  is connected to an Iout output terminal. The collector of the transistor Q 33  is connected to the power source Reg. The base and the collector of the transistor Q 31  are connected to the power source Reg. The transistors Q 30  to Q 35  are NPN-type transistors, and form a multiplication and division circuit. 
     A value obtained by dividing a result of multiplication of an input current of the Ia input terminal and an input current of the Ib input terminal by an input current of the Ic input terminal is outputted to the Iout output terminal as a current signal. In addition, the maximum output of the Tout output terminal is limited to be less than the current source Itail. 
       FIG. 6A  and  FIG. 6B  are graphs showing examples of waveforms indicating the input/output characteristics of the arithmetic operator  12  provided in the interleaved converter according to Embodiment 1.  FIG. 6A  shows how the ON time signal Ion changes when the error amplification signal VCOMP is changed. The constant current I 10  is inputted to the Ia input terminal and the Ib input terminal of the multiplication and division circuit  122 , and the current signal Ivcomp is inputted to the Ic input terminal of the multiplication and division circuit  122 . The expression for the configuration of the multiplication and division circuit  122  is (current source I 10 )×(current source I 10 )/(current signal Ivcomp). Accordingly, the ON time signal Ion has such a characteristic as to be inversely proportional to the error amplification signal VCOMP. 
       FIG. 6B  shows how the OFF time signal Ioff changes when the midpoint voltage VIN of the first voltage divider resistor is changed with the midpoint voltage VFB of the second voltage divider resistor being maintained at a constant value. The ON time signal Ion and an input/output difference signal (Ivfb−Ivin) are inputted to the Ia input terminal and the Ib input terminal of the multiplication and division circuit  123 . Here, the input/output difference signal (Ivfb−Ivin) is obtained by subtracting the current signal Ivin proportional to the midpoint voltage VIN of the first voltage divider resistor from the current signal Ivfb proportional to the midpoint voltage VFB of the second voltage divider resistor. The current signal Ivin proportional to the midpoint voltage VIN of the first voltage divider resistor is inputted to the Ic input terminal of the multiplication and division circuit  123 . 
     The expression for the configuration of the multiplication and division circuit  123  is ON time signal Ion×input/output difference signal (Ivfb−Ivin)/(current signal Ivin proportional to the midpoint voltage VIN of the first voltage divider resistor). This expression is equivalent to that shown in the expression (7), so that the output of the multiplication and division circuit  123  becomes the OFF time signal Ioff. 
       FIG. 7  is a circuit configuration diagram showing the phase signal generator  13  provided in the interleaved converter according to Embodiment 1. In  FIG. 7 , an Ion input terminal is connected to the base and the collector of a transistor Q 40  as well as the base of a transistor Q 41 . The emitters of the transistors Q 40  and Q 41  are connected to the power source Reg. The collector of the transistor Q 41  is connected to the source of a MOSFET Q 44 . The drain of the MOSFET Q 44  is connected to the drain of a MOSFET Q 46 , one terminal of an oscillator capacitor C 1 , a non-inverting input terminal of a comparator CP 1 , and an inverting input terminal of a comparator CP 2 . The gate of the MOSFET Q 44  is connected to the gate of the MOSFET Q 46  and an inverted output Qb of an RS flip-flop FF 1 . 
     The source of the MOSFET Q 46  is connected to the collector of a transistor Q 43 . The base of the transistor Q 43  is connected to the base and the collector of a transistor Q 42  as well as an Ioff input terminal. The emitters of the transistors Q 42  and Q 43  are grounded. The other terminal of the oscillator capacitor C 1  is grounded. 
     An inverting input terminal of the comparator CP 1  is connected to a terminal for the error amplification signal VCOMP. An output terminal of the comparator CP 1  is connected to a reset terminal of the RS flip-flop FF 1 . A non-inverting input terminal of the comparator CP 2  is connected to one terminal of a reference power source Vref. An output terminal of the comparator CP 2  is connected to a set terminal of the RS flip-flop FF 1 . 
     An output Q of the RS flip-flop FF 1  is connected to an input terminal of a frequency divider circuit  132 . The phase signal Ph 1  and the phase signal Ph 2  are outputted from output terminals of the frequency divider circuit  132 , respectively. The transistor Q 40  and the transistor Q 41  form a second current mirror circuit  131 . The transistor Q 42  and the transistor Q 43  form a third current mirror circuit. The MOSFET Q 44  and the MOSFET Q 46  form a switching circuit. 
       FIG. 8  is a circuit configuration diagram showing the frequency divider circuit  132  in the phase signal generator  13  shown in  FIG. 7 . In  FIG. 8 , an input terminal IN 1  is connected to a T input of a T flip-flop FF 2 . An output Q of the T flip-flop FF 2  is connected to an input terminal of an inverter INV 1 , one of input terminals of an AND circuit AND 2 , an input of a delay circuit DL 1 , and one of input terminals of an exclusive OR circuit EOR 1 . An output terminal of the inverter INV 1  is connected to one of input terminals of an AND circuit AND 1 . An output terminal of the delay circuit DL 1  is connected to the other one of the input terminals of the exclusive OR circuit EOR 1 . An output terminal of the exclusive OR circuit EOR 1  is connected to the other one of the input terminals of the AND circuit AND 2  and the other one of the input terminals of the AND circuit AND 1 . The first phase signal Ph 1  is outputted from an output terminal of the AND circuit AND 1 , and the second phase signal Ph 2  is outputted from an output terminal of the AND circuit AND 2 . 
     The T flip-flop FF 2 , the inverter INV 1 , the exclusive OR circuit EOR 1 , the delay circuit DL 1 , and the AND circuits AND 1  and AND 2  form a frequency divider circuit. 
       FIG. 9  is a chart showing the operation waveforms of the respective components of the phase signal generator  13  shown in  FIG. 7 . In  FIG. 9 , VCOMP represents the output voltage of the error amplifier  11 , Vc 1  the voltage across the oscillator capacitor C 1 , Vref the output voltage of the reference power source Vref, CP 1  the output signal of the comparator CP 1 , CP 2  the output signal of the comparator CP 2 , FF 1 Q the output Q of the RS flip-flop FF 1 , FF 2 Q the output Q of the T flip-flop FF 2 , EOR 1  the output of the exclusive OR circuit EOR 1 , Ph 1  the first phase signal Ph 1 , which is the output of the AND circuit AND 1 , Ph 2  the second phase signal Ph 2 , which is the output of the AND circuit AND 2 . 
     First of all, the ON time signal Ion and the OFF time signal Ioff generated by the arithmetic operator  12  are inputted to the phase signal generator  13 . The ON time signal Ion and the OFF time signal Ioff thus inputted are sent to the oscillator capacitor C 1  through the second current mirror circuit Q 40  and Q 41 , the third current mirror circuit Q 42  and Q 43 , and the switching circuit Q 44  and Q 46 . 
     The switching circuit Q 44  and Q 46  switches, in accordance with the state of the RS flip-flop FF 1 , between the charging of the oscillator capacitor C 1  according to the ON time signal Ion and the discharging of the oscillator capacitor C 1  according to the OFF time signal Ioff. 
     When the RS flip-flop FF 1  is in a set state, the inverted output Qb of the RS flip-flop FF 1  is “L”. At this time, the MOSFET Q 44  of the switching circuit is in an ON state, and the MOSFET Q 46  thereof is in an OFF state. Accordingly, the oscillator capacitor C 1  is charged through the transistors Q 40  and Q 41  in accordance with the ON time signal Ion, so that the voltage Vc 1  across the oscillator capacitor C 1  increases. Once the voltage Vc 1  across the oscillator capacitor C 1  is charged to be equal to or higher than the error amplification signal VCOMP, the output of the comparator CP 1  switches from “L” to “H”, so that the RS flip-flop FF 1  is reset. 
     When the RS flip-flop FF 1  is reset, the inverted output Qb of the RS flip-flop FF 1  is switched to “H”. At this time, the MOSFET Q 44  of the switching circuit is turned OFF, and the MOSFET Q 46  thereof is turned ON. Accordingly, the electric charge accumulated in the oscillator capacitor C 1  is discharged through the transistors Q 42  and Q 43  in accordance with the OFF time signal Ioff, so that the voltage Vc 1  across the oscillator capacitor C 1  decreases. 
     Once the voltage Vc 1  across the oscillator capacitor C 1  is discharged to be equal to or lower than the reference voltage Vref, the output of the comparator CP 2  is switched from “L” to “H”, so that the RS flip-flop FF 1  is set again. The above-described operation is repeated to generate a pulse train, which is then inputted to the frequency divider circuit  132 . 
     The pulse train inputted to the frequency divider circuit  132  is divided in frequency by 2 by the T flip-flop FF 2 . The signal thus divided by the T flip-flop FF 2  becomes a pulse train having a duty ratio of approximately 50% of that of the original inputted pulse train if the frequency of the inputted pulse train does not change to a large extent. On the basis of the pulse train having a duty ratio of 50%, signals synchronized respectively with each rising edge and each falling edge are generated by the delay circuit DL 1 , the inverter INV 1 , the exclusive OR circuit EOR 1 , and the AND circuits AND 1  and AND 2 , and are then outputted as the first phase signal Ph 1  and the second phase signal Ph 2 , respectively. 
     The phase signal generator  13  shown in  FIG. 7  generates phase signals for controlling two converters. For this reason, the ON time signal Ion and the OFF time signal Ioff are each doubled in frequency or the oscillator capacitor C 1  is adjusted to have a half value so that the frequency of charge/discharge of the oscillator capacitor C 1  should be twice the original oscillation frequency. 
       FIG. 10  is a circuit configuration diagram showing the pulse generator  14  provided in the interleaved converter according to Embodiment 1. In  FIG. 10 , an input terminal for the ON time signal Ion is connected to the base and the collector of a transistor Q 50 , the base of a transistor Q 51 , and the base of a transistor Q 52 . The emitters of the transistors Q 50 , Q 51 , and Q 52  are connected to the power source Reg. The collector of the transistor Q 51  is connected to one terminal of a first ON-time generator capacitor C 2 , the drain of a MOSFET Q 54 , and a non-inverting input terminal of a comparator CP 4 . 
     The collector of the transistor Q 52  is connected to one terminal of a second ON-time generator capacitor C 3 , the drain of a MOSFET Q 53 , and a non-inverting input terminal of a comparator CP 3 . 
     An input terminal for the error amplification signal VCOMP is connected to an inverting input terminal of the comparator CP 3  and an inverting input terminal of the comparator CP 4 . The other terminal of the first ON-time generator capacitor C 2  is grounded. An output terminal of the comparator CP 4  is connected to a reset terminal of an RS flip-flop FF 4 . An input terminal for the first phase signal Ph 1  is connected to the gate of the MOSFET Q 54  and a set terminal of the RS flip-flop FF 4 . The source of the MOSFET Q 54  is grounded. An output Q of the RS flip-flop FF 4  is connected to an output terminal for the PWM 1 . 
     The other terminal of the second ON-time generator capacitor C 3  is grounded. An output terminal of the comparator CP 3  is connected to a reset terminal of an RS flip-flop FF 3 . An input terminal for the second phase signal Ph 2  is connected to the gate of the MOSFET Q 53  and a set terminal of the RS flip-flop FF 3 . The source of the MOSFET Q 53  is grounded. An output Q of the RS flip-flop FF 3  is connected to an output terminal for the PWM 2 . 
     The comparator CP 4 , the first ON-time generator capacitor C 2 , the MOSFET Q 54 , and the RS flip-flop FF 4  form a first ON-time generator circuit. The comparator CP 3 , the second ON-time generator capacitor C 3 , the MOSFET Q 53 , and the RS flip-flop FF 3  form a second ON-time generator circuit. 
       FIG. 11  is a chart showing the operation waveforms of the respective components of the pulse generator  14  shown in  FIG. 10 . In  FIG. 11 , VCOMP represents the error amplification signal VCOMP, Vc 1  the voltage across the oscillator capacitor C 1 , Vref the voltage across the reference power source Vref, Ph 1  the first phase signal Ph 1 , Ph 2  the second phase signal Ph 2 , Vc 2  the voltage across the first ON-time generator capacitor C 2 , Vc 3  the voltage across the second ON-time generator capacitor C 3 , PWM 1  the first pulse-train signal, PWM 2  the second pulse-train signal. 
     The ON time signal Ion inputted to the first ON-time generator circuit (or the second ON-time generator circuit) charges the first ON-time generator capacitor C 2  (or the second ON-time generator capacitor C 3 ) through the current mirror circuit formed of the transistors Q 50 , Q 51 , and Q 52 . Once the first phase signal Ph 1  (or the second phase signal Ph 2 ) is inputted to the first ON-time generator circuit (or the second ON-time generator circuit), the MOSFET Q 54  (or the MOSFET Q 53 ) is turned ON, so that the electric charge accumulated in the first ON-time generator capacitor C 2  (or the second ON-time generator capacitor C 3 ) is discharged and the RS flip-flop FF 4  (or the RS flip-flop FF 3 ) is set. 
     The first ON-time generator capacitor C 2  (or the second ON-time generator capacitor C 3 ) is charged in accordance with the ON time signal Ion. Once the voltages Vc 1  and Vc 2  are increased to be equal to or higher than the error amplification signal VCOMP, the output of the comparator CP 4  (or the comparator CP 3 ) is switched to “H”. When the output of the comparator CP 4  (or the comparator CP 3 ) is switched to “H”, the RS flip-flop FF 4  (or the RS flip-flop FF 3 ) is reset. 
     Even after the RS flip-flop FF 4  (or the RS flip-flop FF 3 ) is reset, the first ON-time generator capacitor C 2  (or the second ON-time generator capacitor C 3 ) is continuously charged, so that the voltage across the first ON-time generator capacitor C 2  (or the second ON-time generator capacitor C 3 ) increases. Once the first phase signal Ph 1  (or the second phase signal Ph 2 ) is inputted, the MOSFET Q 54  (or the MOSFET Q 53 ) is turned ON, so that the charge accumulated in the first ON-time generator capacitor C 2  (or the second ON-time generator capacitor C 3 ) is discharged and the RS flip-flop FF 4  (or the RS flip-flop FF 3 ) is set again. With the above-described operation repeated, the pulse generator  14  generates a pulse train. 
     On the basis of the pulse trains having mutually different phases which are generated by the pulse generator  14 , the drive circuit  15  generates drive signals for driving the switching elements Q 1  and Q 2 , and thereby drives the switching elements Q 1  and Q 2 . 
     As described so far, since the phase signals are obtained from the result of arithmetic operation on the input voltage, the output voltage, and the error amplification signal, favorable phase signals can be generated even immediately after the start of operation. In addition, a zero current state can be detected with no detecting of currents flowing through the reactors L 1  and L 2 . For this reason, it is possible to suppress, as much as possible, the complicating of a circuit due to an increase in the number of converters connected in parallel without increasing the number of peripheral components such as auxiliary winding. As a result, an inexpensive interleaved converter can be provided. 
     Embodiment 2 
       FIG. 12  is a circuit configuration diagram showing an interleaved converter according to Embodiment 2 of the present invention. In  FIG. 12 , a first series circuit, which includes a switching element Q 1  made of a MOSFET and a rectifier D 3  made of a free-wheeling diode, is connected to both terminals of an input power source Vin formed of a DC power source. A step-down reactor L 3  has one terminal connected to the cathode of the rectifier D 3  and the other terminal grounded through a smoothing capacitor Co. 
     A second series circuit, which includes a switching element Q 2  made of a MOSFET and a rectifier D 4  made of a free-wheeling diode, is connected to both terminals of the input power source Vin. A step-down reactor L 4  has one terminal connected to the cathode of the rectifier D 4  and the other terminal grounded through the smoothing capacitor Co. 
     A third series circuit, which includes a switching element Q 3  made of a MOSFET and a rectifier D 5  made of a free-wheeling diode, is connected to both terminals of the input power source Vin. A step-down reactor L 5  has one terminal connected to the cathode of the rectifier D 5  and the other terminal grounded through the smoothing capacitor Co. 
     A first voltage divider resistor formed of a resistor R 1  and a resistor R 2  is connected to both terminals of the input power source Vin. A second voltage divider resistor formed of a resistor R 3  and a resistor R 4  is connected to both terminals of the smoothing capacitor Co. A control circuit  10   a  generates and outputs gate drive signals for the switching elements Q 1 , Q 2 , and Q 3  based on a midpoint voltage VIN of the first voltage divider resistor and a midpoint voltage VFB of the second voltage divider resistor. 
     The control circuit  10   a  includes an error amplifier  11 , an arithmetic operator  12   a , a phase signal generator  13   a , a pulse generator  14   a , and a drive circuit  15   a . The arithmetic operator  12   a  is configured to receive the midpoint voltage VIN of the first voltage divider resistor, the midpoint voltage VFB of the second voltage divider resistor, and an error amplification signal VCOMP from the error amplifier  11 , and perform arithmetic operation on these voltages. With the output of the arithmetic operation, the arithmetic operator  12   a  generates and outputs an ON time signal Ion and an OFF time signal Ioff. The ON time signal Ion is a signal proportional to an ON time of the switching elements Q 1 , Q 2 , and Q 3 , and the OFF time signal Ioff is a signal proportional to an OFF time of the switching elements Q 1 , Q 2 , and Q 3 . The phase signal generator  13   a  is configured to generate and output a first phase signal Ph 1 , a second phase signal Ph 2 , and a third phase signal Ph 3 , based on the ON time signal Ion and the OFF time signal Ioff. 
     The pulse generator  14   a  is configured to generate and output a first pulse-train signal PWM 1 , a second pulse-train signal PWM 2 , and a third pulse-train signal PWM 3  having the same duty ratio and mutually different phases, based on the first phase signal Ph 1 , the second phase signal Ph 2 , the third phase signal Ph 3 , and the ON time signal Ion. The drive circuit  15   a  is configured to generate and output a first drive signal Vdr 1  for driving the switching element Q 1 , a second drive signal Vdr 2  for driving the switching element Q 2 , and a third drive signal Vdr 3  for driving the switching element Q 3 , based on the first pulse-train signal PWM 1 , the second pulse-train signal PWM 2 , and the third pulse-train signal PWM 3   
     The switching element Q 1 , the rectifier D 3 , and the step-down reactor L 3  form a first converter. The switching element Q 2 , the rectifier D 4 , and the step-down reactor L 4  form a second converter. The switching element Q 3 , the rectifier D 5 , and the step-down reactor L 5  form a third converter. The first converter, the second converter, and the third converter are connected to one another at the respective input terminals as well as at the respective output terminals, thereby forming a buck interleaved converter. 
     The buck converter is configured to output an output voltage Vo that is lower than an input voltage Vin, in accordance with the ON/OFF operations of the switching elements. When the switching element Q 1  (or Q 2  or Q 3 ) is ON, current flows from Vin, through Q 1  (or Q 2  or Q 3 ), L 3  (or L 4  or L 5 ), and Co, to Vin in this order, so that an energy of the magnetic flux is accumulated in the step-down reactor L 3  (or L 4  or L 5 ) and concurrently electric charge is accumulated in the smoothing capacitor Co. 
     When the switching element Q 1  (or Q 2  or Q 3 ) is OFF, current flows from L 3  (or L 4  or L 5 ), through Co and D 3  (or D 4  or D 5 ), to L 3  (or L 4  or L 5 ) in this order, so that the energy of the magnetic flux accumulated in the step-down reactor L 3  (or L 4  or L 5 ) is released. This operation is expressed by the following expression: 
     
       
         
           
             
               
                 
                   
                     Δ 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     IL 
                   
                   = 
                   
                     
                        
                       
                         
                           
                             Vin 
                             - 
                             Vo 
                           
                           L 
                         
                         · 
                         Ton 
                       
                        
                     
                     ≤ 
                     
                       
                          
                         
                           
                             
                               Vo 
                               + 
                               VF 
                             
                             L 
                           
                           · 
                           Toff 
                         
                          
                       
                       . 
                     
                   
                 
               
               
                 
                   Λ 
                   ⁡ 
                   
                     ( 
                     8 
                     ) 
                   
                 
               
             
           
         
       
     
     In the expression (8), ΔIL represents an amount of change in current flowing through the step-down reactor L 3  (or L 4  or L 5 ), Vin the voltage across the input power source Vin, Vo the voltage across the smoothing capacitor Co, VF the forward drop voltage of the rectifier D 3  (or D 4  or D 5 ), L the inductance value of the step-down reactor L 3  (or L 4  or L 5 ), Ton the ON time of the switching element Q 1  (or Q 2  or Q 3 ), Toff the OFF time of the switching element Q 1  (or Q 2  or Q 3 ). 
     If the OFF time Toff in the expression (8) is obtained, the following expression (9) is obtained: 
     
       
         
           
             
               
                 
                   Toff 
                   = 
                   
                     
                       
                         Vin 
                         - 
                         Vo 
                       
                       
                         Vo 
                         + 
                         VF 
                       
                     
                     · 
                     
                       Ton 
                       . 
                     
                   
                 
               
               
                 
                   Λ 
                   ⁡ 
                   
                     ( 
                     9 
                     ) 
                   
                 
               
             
           
         
       
     
     Accordingly, as in the case of Embodiment 1, it is possible to obtain the ON time Ton and the OFF time Toff of the switching elements Q 1  to Q 3 , or the ON time signal Ion proportional to the ON time and the OFF time signal Ioff proportional to the OFF time, by means of the arithmetic operator  12   a.    
       FIG. 13  is a circuit configuration diagram showing the arithmetic operator  12   a  provided in the interleaved converter according to Embodiment 2. The arithmetic operator  12   a  shown in  FIG. 13  has a configuration formed by adding a resistor Ron and a resistor Roff to the arithmetic operator  12  shown in  FIG. 4 . The resistor Ron has one terminal connected to an output terminal for the ON time signal Ion and the other terminal grounded. The resistor Roff has one terminal connected to an output terminal for the OFF time signal Ioff and the other terminal grounded. 
     The resistor Ron and the resistor Roff are configured to convert current signals into voltage signals for the ON time signal Ion and the OFF time signal Ioff obtained by the arithmetic operation of the arithmetic operator  12   a , respectively. 
       FIG. 14  is a circuit configuration diagram showing the phase signal generator  13   a  provided in the interleaved converter according to Embodiment 2. In the phase signal generator  13   a  shown in  FIG. 14 , a constant current source I 61  has one terminal connected to the power source Reg and the other terminal grounded through an oscillator capacitor C 5 . A comparator CP 1  has an inverting input terminal connected to the ON time signal Ion and a non-inverting input terminal connected to a connection point of the constant current source I 61  and the oscillator capacitor C 5 . 
     A constant current source I 60  has one terminal connected to the power source Reg and the other terminal grounded through an oscillator capacitor C 4 . A comparator CP 2  has an inverting input terminal connected to the OFF time signal Ioff and a non-inverting input terminal connected to a connection point of the constant current source I 60  and the oscillator capacitor C 4 . An RS flip-flop FF 1  has a set terminal connected to an output terminal of the comparator CP 2  and a reset terminal connected to an output terminal of the comparator CP 1 . 
     A MOSFET Q 60  has the drain connected to a connection point of the constant current source I 61  and the oscillator capacitor C 5 , the source grounded, and the gate connected to an inverted output Qb of the RS flip-flop FF 1 . A MOSFET Q 61  has the drain connected to a connection point of the constant current source I 60  and the oscillator capacitor C 4 , the source grounded, and the gate connected to an output Q of the RS flip-flop FF 1 . A frequency divider circuit  132   a  is configured to receive the output Q of the RS flip-flop FF 1  and output phase signals Ph 1 , Ph 2 , and Ph 3 . 
       FIG. 15  is a circuit configuration diagram showing the frequency divider circuit  132   a  in the phase signal generator  13   a  shown in  FIG. 14 . In the frequency divider circuit  132   a  shown in  FIG. 15 , a T flip-flop FF 2  connected to an input terminal IN 1 , a T flip-flop FF 2   a , and an AND circuit AND 4  form a ternary counter. An output Q 0  of the T flip-flop FF 2 , which corresponds to a least significant bit output of the ternary counter, is connected to an input terminal of an AND circuit AND 4  and an input terminal of an AND circuit AND 3 . 
     An output Q 1   a  of the T flip-flop FF 2   a , which corresponds to a most significant bit output of the ternary counter, is connected to the other input terminal of the AND circuit AND 4 , an input terminal of a delay circuit DL 1 , an input terminal of an AND circuit AND 5 , and an input terminal of an exclusive OR circuit EOR 1 . An output terminal of the delay circuit DL 1  is connected to the other input terminal of the exclusive OR circuit EOR 1 . An output terminal of the exclusive OR circuit EOR 1  is connected to the other input terminal of the AND circuit AND 5 . The third phase signal Ph 3  is outputted from an output terminal of the AND circuit AND 5 . 
     An output terminal of the AND circuit AND 3  is connected to an input terminal of a delay circuit DL 2 , an input terminal of an AND circuit AND 6 , and an input terminal of an exclusive OR circuit EOR 2 . An output terminal of the delay circuit DL 2  is connected to the other input terminal of the exclusive OR circuit EOR 2 . An output terminal of the exclusive OR circuit EOR 2  is connected to the other input terminal of the AND circuit AND 6 . The second phase signal Ph 2  is outputted from an output terminal of the AND circuit AND 6 . 
     An inverted output Qb of the T flip-flop FF 2   a , which is an inverted signal of the most significant bit of the ternary counter, is connected to an input terminal of a delay circuit DL 3 , an input terminal of an AND circuit AND 7 , and an input terminal of an exclusive OR circuit EOR 3 . An output terminal of the delay circuit DL 3  is connected to the other input terminal of the exclusive OR circuit EOR 3 . An output terminal of the exclusive OR circuit EOR 3  is connected to the other input terminal of the AND circuit AND 7 . The first phase signal Ph 1  is outputted from an output terminal of the AND circuit AND 7 . 
       FIG. 17  is a circuit configuration diagram showing the pulse generator  14   a  provided in the interleaved converter according to Embodiment 2. In  FIG. 17 , an input terminal for the ON time signal Ion is connected to inverting input terminals of comparators CP 3 , CP 4 , and CP 5 . A terminal of each of constant current sources  120 ,  121 , and  122  is connected to the power source Reg. 
     The other terminal of the constant current source I 22  is connected to one terminal of a first ON-time generator capacitor C 11 , the drain of a MOSFET Q 53 , and a non-inverting input terminal of the comparator CP 3 . The other terminal of the first ON-time generator capacitor C 11  and the source of the MOSFET Q 53  are grounded. The other terminal of the constant current source I 21  is connected to one terminal of a second ON-time generator capacitor C 12 , the drain of a MOSFET Q 54 , and a non-inverting input terminal of the comparator CP 4 . The other terminal of the second ON-time generator capacitor C 12  and the source of the MOSFET Q 54  are grounded. The other terminal of the constant current source I 20  is connected to one terminal of a third ON-time generator capacitor C 13 , the drain of a MOSFET Q 55 , and a non-inverting input terminal of the comparator CP 5 . The other terminal of the third ON-time generator capacitor C 13  and the source of the MOSFET Q 55  are grounded. 
     An output terminal of the comparator CP 3  is connected to a reset terminal of an RS flip-flop FF 3 . An input terminal for the first phase signal Ph 1  is connected to a set terminal of the RS flip-flop FF 3 . An output Q of the RS flip-flop FF 3  is connected to an output terminal for the PWM 1 . An inverted output Qb of the RS flip-flop FF 3  is connected to the gate of the MOSFET Q 53 . 
     An output terminal of the comparator CP 4  is connected to a reset terminal of an RS flip-flop FF 4 . An input terminal for the second phase signal Ph 2  is connected to a set terminal of the RS flip-flop FF 4 . An output Q of the RS flip-flop FF 4  is connected to an output terminal for the PWM 2 . An inverted output Qb of the RS flip-flop FF 4  is connected to the gate of the MOSFET Q 54 . 
     An output terminal of the comparator CP 5  is connected to a reset terminal of an RS flip-flop FF 5 . An input terminal for the third phase signal Ph 3  is connected to a set terminal of the RS flip-flop FF 5 . An output Q of the RS flip-flop FF 5  is connected to an output terminal for the PWM 3 . An inverted output Qb of the RS flip-flop FF 5  is connected to the gate of the MOSFET Q 55 . 
     The comparator CP 3 , the first ON-time generator capacitor C 11 , the MOSFET Q 53 , and the RS flip-flop FF 3  form a first ON-time generator circuit. The comparator CP 4 , the second ON-time generator capacitor C 12 , the MOSFET Q 54 , and the RS flip-flop FF 4  form a second ON-time generator circuit. The comparator CP 5 , the third ON-time generator capacitor C 13 , the MOSFET Q 55 , and the RS flip-flop FF 5  form a third ON-time generator circuit. The phase signal generator  13   a  is configured to generate phase signals for controlling three comparators. For this reason, the current values of the constant current sources  160  and  161  are adjusted or each of the oscillator capacitors C 4  and C 5  is adjusted to have an one-third value so that the frequency of charge/discharge of each of the oscillator capacitors C 4  and C 5  should be three times as the original oscillation frequency. 
       FIG. 16  is a chart showing the operation waveforms of the respective components of the phase signal generator  13   a . In  FIG. 16 , Ion represents the ON time signal Ion, Vc 4  the voltage across the oscillator capacitor C 4 , Ioff the OFF time signal Ioff, Vc 5  the voltage across the oscillator capacitor C 5 , CP 1  the output signal of the comparator CP 1 , CP 2  the output signal of the comparator CP 2 , FF 2 Q the output signal of the RS flip-flop FF 2 , Q 0  the least significant bit output of the ternary counter, Q 1   a  the most significant bit output of the ternary counter, Ph 1  the first phase signal, Ph 2  the second phase signal, Ph 3 , the third phase signal, Vc 11  the voltage across the first ON-time generator capacitor C 11 , Vc 12  the voltage across the second ON-time generator capacitor C 12 , Vc 13  the voltage across the third ON-time generator capacitor C 13 , PWM 1  the first pulse-train signal, PWM 2  the second pulse-train signal, PWM 3  the third pulse-train signal. 
     Next, the operation of the phase signal generator  13   a  will be described with reference to  FIG. 16 . First, when the RS flip-flop FF 1  is in a set state, the MOSFET Q 61  is ON and the MOSFET Q 60  is OFF. Since the MOSFET Q 61  is ON, the oscillator capacitor C 4  is discharged. On the other hand, since the MOSFET Q 60  is OFF, the oscillator capacitor C 5  is charged by the constant current source I 61 , so that the voltage across the oscillator capacitor C 5  increases. 
     Once the voltage across the oscillator capacitor C 5  is increased to be equal to or higher than the potential of the ON time signal Ion, the output of the comparator CP 1  is switched from “L” to “H”, so that the RS flip-flop FF 1  is reset. When the RS flip-flop FF 1  is reset, the MOSFET Q 60  is turned ON, so that the oscillator capacitor C 5  is discharged and the MOSFET Q 61  is turned OFF. 
     When the MOSFET Q 61  is turned OFF, the oscillator capacitor C 4  is charged by the constant current source I 60 . Once the voltage across the oscillator capacitor C 4  is increased to be equal to the potential of the OFF time signal Ioff, the comparator CP 2  is switched from “L” to “H”, so that the RS flip-flop FF 1  is set. The above-described operation is repeated, so that a pulse-train signal is outputted from the output terminal of the RS flip-flop FF 1 . The pulse-train signal varies in the ratio and frequencies of the “H” level and the “L” level depending on the ON time signal Ion and the OFF time signal Ioff, respectively. 
     When inputted to the frequency divider circuit  132   a , the output signal of the RS flip-flop FF 1  is divided by 3 by the ternary counter, thereby being converted into a digital signal having the most significant bit Q 1   a  and the least significant bit Q 0 . The digital signal thus obtained through the conversion is converted into the phase signals Ph 1 , Ph 2 , Ph 3 , having phases mutually shifted by approximately 120 degrees, by the frequency divider circuit  132   a  formed of the AND circuits ANDS to AND 7 , the exclusive OR circuits EOR 1  to EOR 3 , and the delay circuits DL 1  to DL 3 . 
     Each of the phase signals Ph 1 , Ph 2 , and Ph 3  is inputted to the pulse generator  14   a  shown in  FIG. 17 , so that the pulse-train signals PWM 1 , PWM 2 , and PWM 3  having mutually different phases are generated. The pulse-train signals PWM 1 , PWM 2 , and PWM 3  are converted respectively into the drive signals Vdr 1 , Vdr 2 , and Vdr 3  for driving the switching elements Q 1  to Q 3  by the drive circuit  15   a . As a result, the interleaved converter is operated. 
     As described so far, the interleaved converter according to Embodiment 2 provides the same advantageous effects as those provided by the interleaved converter according to Embodiment 1. 
     It should be noted that the present invention is not limited to Embodiments 1 and 2 described so far. Although the analogue arithmetic operators are used as the arithmetic operators  12  and  12   a  in Embodiments 1 and 2, digital arithmetic operators may be used instead. In addition, the converter circuits may be buck boost converters instead of the boost converters and the buck converters. Alternatively, the converter circuits may be forward converters, flyback converters, or resonant converters. Moreover, although the forward step-down voltages of the rectifiers D 1  to D 5  are omitted in the arithmetic operators  12  and  12   a , correction signals may be added in consideration of the forward step-down voltages. 
     Furthermore, even if the number of converters connected in parallel is increased, the present invention may be adapted by setting the oscillation frequency of each oscillator capacitor as a high frequency suitable for the number of converters and by increasing the number by which the frequency is divided by the frequency divider circuit. 
     According to the present invention, since the phase signal is obtained from the result of arithmetic operation on the input voltage, the output voltage, and the error amplification signal, favorable phase signals can be generated immediately after the start of operation. In addition, a zero current state can be detected with no detecting of currents flowing through the reactors. For this reason, it is possible to suppress, as much as possible, the complicating of a circuit due to an increase in the number of converters connected in parallel without increasing the number of peripheral components such as auxiliary winding. As a result, an inexpensive interleaved converter can be provided. 
     The present invention may be employed as a control system for the case where multiple converters are connected in parallel and controlled with mutually shifted phases.