Abstract:
A microwave radio frequency bidirectional energy flow-capable antenna method and related antenna of the physically conformal microstrip transmission line, traveling wave and leaky wave characterizations; the antenna is especially suited to vehicle mounting. The disclosed antenna operates in an EH 1  or other above dominant mode energy wave propagation configuration, a configuration at least partially achieved by an array of selected-location radiating element shortings to an antenna-underlying transmission line ground plane element. Comparisons of the disclosed antenna with characteristics of a similarly classified antenna of somewhat lesser desirable but know characteristics are included.

Description:
RIGHTS OF THE GOVERNMENT 
   The invention described herein may be manufactured and used by or for the Government of the United States for all governmental purposes without the payment of any royalty. 
   BACKGROUND OF THE INVENTION 
   One of the “Holy Grails” for antenna engineers working in the aircraft and other vehicle fields, where aerodynamic drag and vehicle profile are important, is achieving an antenna with wide bandwidth, high efficiency, a convenient radiation pattern, and a small addition to vehicle profile. This latter characteristic may be considered, with the use of other words, as a need for a vehicle “conformal antenna.” The need for these characteristics extends significantly into the world of the stealth aircraft since non-conformal protuberances on an airframe provide a substantial radar signal reflection or return point addition to the aircraft&#39;s radar signature. These desired antenna characteristics typically are, however, conflicting in nature and thus an antenna engineer must often make trade-offs amongst these needs. 
   There are a variety of conformal antennas used in microwave signal and aircraft practice, but perhaps the most studied of these is the microstrip patch antenna. As a result of its relative simplicity with respect to both modeling and construction, the patch antenna has been a subject of extensive research and use for over thirty years. Munson [1] performed seminal work on this antenna as did Carver and Mink [2]. (Numbers of this configuration herein refer to entries in the list of references at the close of this specification.) The simple approximate models developed by these authors have been used since their publication and are now included in the subject matter of many Engineering School undergraduate antenna courses [3]. 
   One of the major challenges associated with the patch antenna is however, the relatively narrow bandwidth such an antenna achieves [4]. Such antennas, if probe or microstrip transmission line fed, have a bandwidth of typically less than 5% and often less than 2% [3]. Increasing the substrate thickness used with these antennas can increase this bandwidth; however, surface waves can be excited in such patch antennas and this leads to a rather serious reduction in efficiency. This reduction can be limited by the introduction of shorting pins, or a cavity that have the effect of squelching surface waves. However, care must be used in achieving such surface wave reductions since placement of metal near the radiating edges of a patch antenna has a significant impact on its properties. Moreover since patch antennas are usually used in large arrays, in part because of their low cost and low gain, shorting pins or cavities cannot always be used due to the proximity of the antenna elements to each other. The result is strong surface wave coupling between adjacent antennas and this complicates the antenna synthesis task. Alternative feeding mechanisms can be used to increase the achieved bandwidth, without exciting surface waves; however, the achievable bandwidth is typically on the order of 20% to 60% [5] but certainly bandwidths of 2:1 or 10:1 are not achievable with any manner of feeding a patch antenna. 
   Another approach to increasing patch antenna bandwidth, without a commensurate reduction in efficiency involves the use of magneto-dielectric materials [6] in the antenna. However, the relatively high efficiency that can otherwise be achieved with patch antennas requires low loss magnetic materials. Such materials are difficult to realize at high frequencies, at frequencies greater than 1 gigahertz for example. 
   From another perspective, there are a group of antennas that are inherently of wide bandwidth and have reasonable efficiency. These antennas include printed spirals (including slot spirals), circular log-periodic arrays as well as helix, bicone, and sleeve antennas. A general theory concerning these and other frequency independent antennas has in fact been presented by Rumsey and is described by Thiele [3]. The first two of these wide band antennas are amenable to conformal installation as in an airframe while the latter types typically are protruding antennas. However, like the patch antenna, the radiation pattern for these antennas depends on feed conditions or mode of operation chosen and has a peak normal to the platform in which it is installed. Examples of feed conditions that will result in a pattern peak away from this direction include higher-order mode excitation for the patch or a phase array of elements with the excitation feed phases chosen to steer the beam. However, it is a well-known fact that for a finite array of elements, there are scan limits on the beam for such elements. 
   The antenna of the present invention provides what is believed to be a useful addition, perhaps even a breath of fresh air, to this antenna selection scene. 
   SUMMARY OF THE INVENTION 
   The present invention provides a microwave antenna suited for use as a conformal antenna. 
   It is therefore an object of the present invention to provide a traveling wave antenna that is based on the use of microstrip transmission line-embodied electrical conductors. 
   It is another object of the invention to provide an improved leaky wave antenna. 
   It is another object of the invention to provide physical size improvement for a leaky wave antenna. 
   It is another object of the invention to provide an improved traveling wave form of a leaky wave antenna. 
   It is another object of the invention to provide a leaky wave type of traveling wave antenna in which the antenna conductor is a solid and undisturbed conductor having either of two width dimensions. 
   It is another object of the invention to provide a leaky wave type of traveling wave antenna based upon use of a type of transmission line conductor as the radiating element. 
   It is another object of the invention to provide a transmission line type of leaky wave traveling wave antenna in which the null effects of certain transmission line perturbations are achieved by alternate and preferable arrangements. 
   It is another object of the invention to provide an improved leaky wave type of traveling wave antenna. 
   It is another object of the invention to provide an improved leaky wave type of traveling wave antenna in which the antenna element may have either of two physical width dimensions. 
   It is another object of the invention to provide a leaky wave type of traveling wave antenna in which the antenna conductor is a solid and undisturbed conductor. 
   It is another object of the invention to provide an improved leaky wave type of traveling wave antenna having a basic element configuration that may be repeated in a multiple element array. 
   It is another object of the invention to provide an improved leaky wave type of traveling wave antenna in which the antenna conductor or conductors may be configured in other than straight line shapes. 
   It is another object of the invention to provide an ultra thin traveling wave antenna. 
   It is another object of the invention to provide a traveling wave antenna having high efficiency and an end-fire radiation pattern. 
   It is another object of the invention to provide an antenna making use of a higher order energization and operating mode in a transmission line element. 
   It is another object of the invention to provide a microwave antenna suited for use as a high performance airframe-mounted conformal antenna. 
   It is another object of the invention to provide an improved leaky wave type of traveling wave antenna that may be used for both signal receiving and signal transmitting purposes. 
   It is another object of the invention to provide an improved leaky wave type of traveling wave antenna having a more desirable leakage rate than is achieved by a prior art Menzel antenna. 
   It is another object of the invention to provide an improved leaky wave type of traveling wave antenna in which omission of active element slots used in a related prior art Menzel antenna precludes existence of slot sourced antenna emissions and hence enables lower emission of undesirable cross polarized radiation components. 
   It is another object of the invention to provide an improved leaky wave traveling wave antenna in which use of shorting based suppression of fundamental mode energy propagation in a microstrip transmission line element is advantageous over the slot achieved suppression of fundamental mode energy propagation employed in a related prior art Menzel antenna. 
   It is another object of the invention to provide an improved leaky wave antenna that is easier to feed than previous higher order mode leaky wave antennas. 
   It is another object of the invention to provide an improved leaky wave traveling wave antenna array in which a reduced degree of mutual coupling between array elements is achieved. 
   These and other objects of the invention will become apparent as the description of the representative embodiments proceeds. 
   These and other objects of the invention are achieved by the wideband traveling wave and leaky wave antenna method of communicating microwave radio frequency energy with a vehicle comprising the steps of: 
   disposing an elongated, electrically insulated outside conductor and ground plane inside conductor, microstrip transmission line antenna assembly in a conforming physical relationship with a selected surface portion of said vehicle; 
   energizing said elongated metal antenna element outside conductor portion of said microstrip transmission line antenna assembly in an energy radiating higher order operating mode; 
   suppressing dominant fundamental mode energy propagation along said elongated metal antenna element outside conductor of said microstrip transmission line to achieve an electrical field phase reversal pattern about an orthogonal lengthwise axis of said outside conductor antenna element; 
   said suppressing step including establishing an electrical field null along said lengthwise axis portion of said elongated antenna element by shorting said lengthwise axis portion of said outside conductor antenna element to said ground plane inside conductor of said microstrip transmission line at a plurality of lengthwise axis locations extending along said elongated outside conductor antenna element in locations wherein said dominant fundamental mode tends to be of greatest amplitude when not suppressed and said energy radiating higher order operating mode tends to be of small amplitude with presence of dominant fundamental mode suppression. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The accompanying drawings incorporated in and forming a part of the specification, illustrate several aspects of the present invention and together with the description serve to explain the principles of the invention. In the drawings: 
       FIG. 1  shows a non radiating EH 0  dominant mode electric field distribution in a microstrip transmission line structure. 
       FIG. 2  shows a radiating EH 1  mode electric field distribution in a microstrip transmission line structure. 
       FIG. 3  shows a theoretical prior art Menzel leaky wave antenna. 
       FIG. 4  shows a fabricated prior art Menzel leaky wave antenna. 
       FIG. 5   a  shows details and dimensions for one arrangement of a present invention antenna. 
       FIG. 5   b  shows details and dimensions for a second arrangement of a present invention antenna. 
       FIG. 6  shows complete antenna front side detail of a present invention leaky wave antenna. 
       FIG. 7  shows a slightly enlarged backside view of the  FIG. 6  antenna. 
       FIG. 8  shows a left hand end view of the  FIG. 6  and  FIG. 7  antenna. 
       FIG. 9  shows the  FIG. 7  defined cross-sectional view of the present invention antenna. 
       FIG. 10  shows an elevation view and main beam direction for a present invention leaky wave antenna. 
       FIG. 11  shows one radiation pattern comparison for a present invention antenna. 
       FIG. 12  shows a dipole arrangement for near field measurements of a Menzel or present invention antenna. 
       FIG. 13  shows a monopole arrangement for measurements of a present invention antenna. 
       FIG. 14  a comparison of present invention and Menzel antennas accomplished with a  FIG. 12  type of measurement apparatus. 
       FIG. 15  shows a comparison of present invention and Menzel antennas accomplished with a  FIG. 13  type of measurement apparatus. 
       FIG. 16  shows details of a via structure suitable for use with the present invention printed circuit antenna conductors. 
       FIG. 17  shows a polar radiation pattern for the  FIG. 5   a  antenna. 
       FIG. 18  shows a polar radiation pattern for the  FIG. 5   b  antenna. 
       FIG. 19  shows two curved present invention antennas, one fed on the outside radius and the other on the inside radius. 
       FIG. 20  shows a polar radiation pattern for the  FIG. 19  antenna. 
   

   DETAILED DESCRIPTION 
   The antenna of the present invention is an inherently wide bandwidth antenna belonging to the general class of traveling wave antennas. Such traveling wave antennas also include the Beverage-can antenna and the rhombic antenna for examples. Antennas of this type utilize a load element at the end of the antenna to dampen undesirable back energy wave reflections and thus have a limit on their efficiency since the energy dissipated in this load is not radiated. As these antennas become electrically longer however, the main beam of the antenna desirably squints towards the direction of propagation and this characteristic tends to overcome the load energy loss. An enlightening overview of wide bandwidth antennas is disclosed in reference [3] herein; this reference and each of the other references identified in this document are hereby incorporated by reference herein. 
   The presently desired performance aircraft conformal version of a traveling wave antenna can be implemented using microstrip transmission line technology. The fundamental excitation or operating mode for such a microstrip transmission line of course intentionally does not radiate energy. Such a non-radiating microstrip transmission line arrangement and the related electric and magnetic field patterns are represented in the  FIG. 1  drawing herein. The fundamental excitation or operating mode represented in the  FIG. 1  drawing is, as is known in the art, achieved by carefully considered microstrip energization and by controlling energy wave propagation characteristics along the antenna element itself. Additional information concerning the latter of these mode-favoring techniques may be appreciated from subsequent discussions herein. 
   It is also well known in the antenna art that a microstrip transmission line does radiate if it is excited in its first higher order mode with a suppression of the fundamental or dominant mode fields. Hence, it is feasible to realize a traveling wave antenna using microstrip transmission line if the transmission line and its feed components are properly developed for a first or other higher order operating mode. Such an antenna will in principle have wide bandwidth, a near “end-fire” radiation pattern, high efficiency, and be ultra thin in profile (e.g. a profile much less than one quarter of a wavelength). An antenna of this nature does have the drawback of achieving a radiation pattern peak location or direction that is frequency dependent; however, the impact of this property can be minimized for a range of frequencies given sufficient real estate surrounding the antenna elements as is discussed in reference [9] herein. Peaking characteristics may for example be minimized for a range of frequencies if a tapered configuration is used for the antenna [12]. A new lightweight, low cost, easily fabricated, leaky wave configuration for an antenna of these types is the subject of the present invention. 
   A leaky wave antenna is a special form of traveling wave antenna that is characterized by a wave propagating interior to a guiding structure rather than exterior to the structure as occurs for example in the case of the Beverage-can traveling wave antenna. As seen in  FIG. 1  herein, the dominant operating mode of a standard microstrip transmission line does not radiate radio frequency energy since the guided wave below the upper microstrip conductor is coherent in nature; or, in other words, the guided wave below the upper microstrip conductor is tightly bound to the structure, there being no phase reversal across the upper microstrip conductor. 
   As shown in the drawing of  FIG. 2  herein, when the dominant mode in a standard microstrip transmission line is suppressed, the higher order mode undergoes a phase reversal of the electric field along a centered vertical axis,  200 , and radiation of the first higher order mode occurs. Radiation can occur with the electrical field shown in  FIG. 2  since the E-field lines at the edges of the upper transmission line conductor are in opposite directions allowing the E-field to add in a direction of radiation whereas in  FIG. 1  the E-field lines at the upper conductor edges oppose each other in a direction of radiation. 
   In the presently desired  FIG. 2  related antenna arrangement therefore the guided-wave energy sets up a leaky E field wave exterior to the guiding transmission line structure and thereby “leaks or sheds” power away from the transmission line structure in a controlled manner as the input energy wave propagates from the feed point to the termination point of the transmission line. In doing so, radiation occurs with a peak that squints in the direction of propagation, as is the case with a Beverage-can antenna. In the case of the  FIG. 2  antenna however, the transmission line based antenna is amenable to the conformal installation that is desired for high performance aircraft use. 
   Wolfgang Menzel of Ulm, Germany, proposed in the late 1970&#39;s an interesting example of a leaky wave antenna, a specific antenna that is more fully disclosed in reference [8] herein. The Menzel antenna is also shown in the  FIG. 3  theoretical version and  FIG. 4  fabricated version drawings herein for reference and comparison purposes. The Menzel antenna includes a wide microstrip transmission line having several centerline rectangular slots  300 ,  302  etc. located close to the feed end of the antenna and in the interior of the transmission line conductor. At microwave frequencies these slots create an electric field null, or a virtual ground, along the center  304  of the microstrip conductor causing this portion of the conductor to effectively short to ground. The  FIG. 1  drawing thus shows the relevant field distribution for the center portion of the microstrip conductor of the  FIG. 3  and  FIG. 4  Menzel antenna. This shorting to ground effect in fact allows the first higher order mode of energy propagation along the length of the Menzel microstrip conductor because an electric field null along the microstrip centerline exists. The  FIG. 2  drawing thus shows the relevant EH 1  field lines for the Menzel type of antenna appearing in the  FIG. 3  and  FIG. 4  drawings herein. The  FIG. 4  drawing in fact represents an actually fabricated Menzel antenna  400  using microstrip transmission line and having an inches calibrated comparison measuring scale  402  nearby. H field lines are of course also present in the  FIG. 2  transmission line but are omitted in the interest of drawing clarity. 
   Improvements to the Menzel  FIG. 3  and  FIG. 4  leaky wave microstrip antenna, improvement according to the present invention, are represented in the  FIG. 5  and several subsequent drawings herein. In the  FIG. 5   a  drawing portion of  FIG. 5  there is shown a microstrip transmission line based antenna  500  inclusive of a first one of these improvements. The antenna  500  includes a body portion or radiating element  510 , an input or output electrical energy transmission line segment  511  and a ground plane inclusive electrically insulating substrate member  504 . The phrase “input or output” in this sense refers to the fact that the antennas of the present invention may be used in either or both of the transmitting (i.e., electrical signal to electromagnetic wave transducing) or the receiving (i.e., electromagnetic wave to electrical signal transducing) functions even though it is often convenient to speak or think primarily in terms of the transmitting function in describing the invention. 
   The  FIG. 5   a  antenna further includes the two halves  501  and  502  of the radiating element  510  and a row of electrical connections  503 , intermediate these two halves  501  and  502 , by which the lengthwise extending center portions of the radiating element  510  are multiply connected electrically to a ground plane backside conductor, indicated at  528 , of the  FIG. 5   a  transmission line. The input or output electrical energy transmission line segment  511  includes an enlarged portion  508  acting as an electrical impedance correction or transformer element at microwave frequencies. The impedance corrected portion of the transmission line  511  connects with one corner of the radiating element  510  as is shown at  505  for energy communicating purposes. A ground plane-side received coaxial cable connector is electrically joined with the transmission line  511  as indicated at  512 . The antenna shown in  FIG. 5   a  is tuned for operation in about the 6 to 8 Gigahertz range. The illustrated dimensions can be scaled for use at other operating frequencies. 
   Physical and electrical dimensions for the  FIG. 5   a  antenna appear in the  FIG. 5   a  drawing. At  520  for example is shown the physical dimensions in millimeters desired for the width of the transmission line conductor  511 . Similarly at  516  and  518  in the  FIG. 5  drawing are shown the grounding element pitch and the length dimensions for the  FIG. 5   a  antenna while the length of the transmission line impedance-changing element is indicated at  514  in  FIG. 5 . The electrical length and width dimensions for the  FIG. 5   a  antenna are indicated at  524  and  522  respectively. It is found desirable for the antenna length dimension, L, at  524  to be between 5 and 10 free space wavelengths for the signal being communicated by the antenna  500 . In a similar manner it is found desirable for the antenna width dimension, W, at  522  to be about one third (⅓) of a free space wavelength for the signal being communicated by the antenna. The element identification numbers used in this description of  FIG. 5   a  are re used to the best degree possible in the discussions of ensuing drawings herein in order to maintain a consistent identity for an element once assigned. Newly identified elements in these ensuing drawings are assigned an identification number relating to the drawing-number, generally this identification number bears a factor of 100 relationship to the drawing number. 
   Considering the  FIG. 6  drawing in detail, in this drawing there appears a substrate member  504  that may be fabricated as a printed circuit board having transmission line radiating conductor  510  received thereon and having the  FIG. 7  shown larger grounded plane transmission line conductor  528  received on the backside thereof. The conductors  510  and  528  may be composed of copper, a copper alloy or of other electrically conductive metals including brass or gold. The substrate  504  in  FIG. 6  may be made of a dimensionally stable and high strength material such as Rogers 5870 Duroid PTFE glass fiber or equivalent and may have a thickness of about 0.787 millimeter. This material has a dielectric constant              r  of 2.33 and is available by way of the current World Wide Web address: rogers-corp. Other characteristics including the leakage constant α, propagation constant β and characteristic measurements relating to the  FIG. 6  antenna appear in subsequent paragraphs herein.
   Continuing with describing details of the  FIG. 6  and the related drawings of  FIG. 7 ,  FIG. 8 ,  FIG. 9 , and  FIG. 10  herein, the  FIG. 6  front side view of an antenna according to the present invention also includes the energy conveying transmission line conductor  511  by which transmitter output energy is coupled to the transmission line radiating conductor  510  or received radiation energy is coupled to a radio receiver apparatus. The enlarged portion of the transmission line conductor  511  at  508  serves the function of an impedance matching element in order to provide a characteristic impedance near 50 ohms at the coaxial cable coupling  700  located on the backside surface of the substrate  504  as shown in  FIG. 7 ; connection of this coupling  700  to the transmission line  511  is represented at  512  in the  FIG. 5  and  FIG. 6  drawings and may consist of a soldered connection. A second such coaxial cable coupling  704  appears in the  FIG. 7  drawing and is attached to the radiating conductor  510  as represented at  612  in  FIG. 6 . The couplings  700  and  704  appear in profile view in the  FIG. 8  right end drawing. The gap  608  in the  FIG. 6  view of transmission line conductor  604  allows isolated impedance measuring and other diagnostic measurements of the radiating conductor  510  to be made and is normally absent and replaced with continuation of the conductors  510  and  511  in a completed and serviceable embodiment of the invention, i.e., this gap  608  is normally shorted. 
   One of the above-described conductor  510  to ground plane  528  shorting element conductors is indicated at  616  in the  FIG. 6  drawing and a backside view of this conductor appears at  706  in  FIG. 7 . As indicated by the cutting line  9 — 9  in  FIG. 7  a cross sectional view of the shorting element conductor  616 – 706  appears in the enlarged  FIG. 9  drawing view. As also suggested in this  FIG. 9  view, the illustrated embodiment of the shorting element conductor  616 – 706  may consist of a copper wire segment folded over into adjacency with each of conductors  510  and  528  and then flow-soldered into place. Other ways of achieving the desired conductor  510  to conductor  528  shorting, including the printed circuit via structure shown in  FIG. 16  herein, are of course possible and are considered to be within the scope of the present invention. The number of shorting element conductors  616 – 706  needed in a particular antenna is dependent on the wavelength of the radio frequency energy being considered and is most conveniently expresses as a number of shorting element conductors per wavelength. It is found, for example, that twenty (20) or more shorting elements per wavelength is a satisfactory arrangement for the invention. 
   While considering the via structure shown in the  FIG. 16  drawing it appears appropriate to discuss certain details of this structure as it is usually fabricated in the electrical art. As shown in  FIG. 16  the printed circuit board via  1600  is provided with an aperture  1604  of selected size traversing the electrical insulating material  1602  of the printed circuit board. This aperture  1604  additionally passes through the lower surface conductor  1608  of the printed circuit board and is plated through or otherwise filled with upper surface conductor material  1606  including the material at  1610  that overlaps and thereby makes electrical contact with the lower surface material. A similar overlapping arrangement may be used for connection with the upper surface material  1606  if needed. Drilling, masking and equivalent fabrication procedures may be used to achieve the  FIG. 16  structure and soldering may be used to improve the electrical contact achieved at  1610 . The relatively low electrical impedance and multiple conduction paths achieved by the circular conductor region  1612  in a via is desirable for shorting elements use in the present invention where microwave radio frequencies are involved. 
   According to the  FIG. 5   a  first of the present invention leaky wave microstrip antenna improvements therefore in order to prevent propagation of energy in the EH 0  fundamental mode along the antenna microstrip conductor  510 , closely spaced, ground plane connected, electrical shorting element conductors are disposed along the center line of the conductor  510 . These shorting element conductors may be disposed in the form of the printed circuit board via element shown in the  FIG. 16  drawing herein and may also be grounded metal shunts of the type shown at  614  and  616  in the  FIG. 6  drawing. These shorting element conductors have an effect comparable to the Menzel rectangular slots  300 ,  302  in that they achieve an electric field null in the form of an actual elongated conductor multi point grounding along the center of the microstrip conductor  510 . The physical null thus accomplished in the electric field attending the  FIG. 6  antenna conductor  510  achieves suppression of dominant or EH 0  mode propagation in the conductor  510 , in the manner represented in the  FIG. 1  drawing, and allows propagation of the EH 1  mode in the manner shown in  FIG. 2 . Grounding of the metal shorting elements added for this EH 0  to EH 1  favoring mode change of course means shorting the upper microstrip conductor  510  to the lower or backplane or remaining microstrip conductor,  528  in  FIG. 7 , by way of the numerous added metal shorting elements. We now believe this null achievement through use of radiation element shorting to the ground plane is more effective in suppressing fundamental mode propagation than is the slot achieved null generation used in the Menzel antenna. In addition the grounded metal shorting element of the present invention eliminates the need for transmission line slots that have been found to cause undesirable cross polarized radiation by the Menzel antenna. 
   In addition to achieving the  FIG. 2  EH 1  field pattern, the desirable effect of grounding the center region of antenna transmission line conductor  510  with shorting element conductors also suggests an ability to dispense with half of the antenna conductor  510 , the conductor portion represented by the dotted line  618  in  FIG. 6  and the portion represented at  502  in  FIG. 5 , without detriment to antenna performance. This dispensing is possible because in fact the propagated desired higher order mode has zero amplitude at the location of the vias or shorting conductors whether or not these conductors are present. With this dispensing the resulting antenna element has the appearance shown at  501  in the  FIG. 5   b  drawing. The  FIG. 5   b  drawing thus shows the second of the microstrip antenna improvements contemplated in the present invention. 
   In fact the  FIG. 5   b  width reduction can be achieved without negatively impacting the suppression of fundamental mode propagation in the narrowed conductor  532 . The width dimension of this new transmission line radiating element is shown at  532  in the  FIG. 5   b  drawing and is near ⅙ of a wavelength, i.e., one-half of the width dimension on the  FIG. 5   a  conductor and the related Menzel conductor. As shown at  530  in the  FIG. 5   b  drawing the lower edge of the reduced width conductor is preferably located as close as practical to the  FIG. 16  printed circuit vias or the  FIG. 9  through conductors accomplishing the front conductor to back plane shorting as is possible. The width of the exposed dielectric surface at  534  in the  FIG. 5   b  reduced width antenna is not critical and need only be one half wavelength or more. This new narrow configuration is shown in full conductor length in the fabricated antenna drawing of  FIG. 6 . Since the footprint of the  FIG. 6  antenna is now smaller, an array of such elements for example can be packed closer together with less mutual coupling between elements. 
     FIG. 10  in the drawings shows an elevation view of the antenna in the  FIG. 6  through  FIG. 9  drawings and shows a somewhat exaggerated form of the metal layers of conductors  510  and  528 . Also appearing in  FIG. 10  is an arrow  1000  indicating in general the direction of radiation provided by the present invention antenna as a result of EH1 mode propagation between conductors  510  and  528  and radiation leakage from the outside edges of the conductor  510 . As suggested previously herein, the angle  1002  between a main lobe of the radiation represented by the arrow  1000  and the antenna conductors  602  and  702  is dependent on the length of the radiating conductor  510  and tends to be smaller in size with a longer radiating conductor. More specific details of this and other characteristics of the present invention antenna appear in the  FIG. 11  and subsequent drawings herein. While considering the  FIG. 10  drawing however, it is significant to note that use of a sufficient length of the radiating transmission line conductor  510  is usually an adequate condition for enabling the antenna to radiate about ninety percent of the energy received from a transmission line energy source feeding the antenna. Radiation of this large fraction of the input energy of course also means the amount of energy available for undesirable reflections from the radiating end of the antenna is relatively low and in the ten percent of input energy range. 
   To illustrate performance of the present invention antenna, several measurements comparing a standard Menzel microstrip antenna and the present invention antenna are believed to be informative. A present invention microstrip antenna for measurement and other uses may be created with a state of the art milling machine compatible with the software autoCAD, allowing drawings created in autoCAD to be transferred to accurate tracings of designs etched from copper covered substrate to the accuracy of a tenth of a millimeter. Both the present invention and the Menzel antennas may be fabricated on Rogers 5870 duroid substrate made of PTFE glass fiber with a thickness of 0.787 millimeter. The length of each antenna may be 190 millimeters beginning where the feed transmission line width opens up to the maximum width of the radiating conductor, i.e., beginning at  618  in  FIG. 6 , and ending at the rightmost end of the antenna conductor. The Menzel antenna width is 15 millimeters while the width of the  FIG. 5   b  and  FIG. 7  present invention half-width antenna is 7.5 millimeters for a 6.7 Gigahertz version of the antenna. 
     FIG. 11  in the drawings shows a comparison of main-lobe elevation field strength pattern measurements i.e., half power beam width field strength versus elevation angle for far-field patterns at 6.7 GHz., made with use of the Menzel antenna and with the present invention  FIG. 5   b  antenna in a laboratory setting. These results indicate the present invention antenna notwithstanding its reduced footprint produces radiation pattern similar to that of the Menzel antenna. 
   For leaky wave antennas, it is also desirable to compare antenna performances by way of considering the leakage constants, α, and the phase constants, β. A leakage constant value relates to the pattern beam width and is significant for minimizing the length of the antenna. The phase constant determines the angular location of the pattern peak. From  FIG. 11  it is observed that the HPBW (half-power beam widths) are 16 degrees for the Menzel antenna and 17 degrees for the present invention antenna. This indicates the leakage constant,            , is approximately the same for each antenna. As also seen in  FIG. 11 , the pattern peak is almost at the same angle for the two antennas; this suggests that the phase constant, β, is approximately identical for the two antennas.
   Since far-field characteristics as in  FIG. 11  are but a coarse indicator of the actual source distribution, it is desirable to compare the actual α and β for the two antennas. To accomplish this, measurements of the source distribution may be taken by probing the fields near the antenna in a near-field anechoic chamber adapted for this purpose. Two different probe configurations may be used. One configuration is a resonant dipole as represented in  FIG. 12  and the other a monopole probe as represented in  FIG. 13 . Both of these measurements are useful for determining β, but the results using the dipole are sensitive to dipole height above the tested antenna element making the determination of             difficult. If the test dipole is too close to the antenna, the dipole perturbs the field in an unacceptable manner while the propagating mode requires the probe to be near the antenna. In each probe case, measurements may be taken at increments of one wavelength from one to four wavelengths in total distance. The monopole probe appears most effective in obtaining accurate amplitude distribution results at a distance of greater than 1 wavelength from the antenna under test.
   Results obtained with the two probes are shown in  FIGS. 14 and 15  of the drawings for the dipole and monopole probes, respectively. As is evident, in these drawings the electric fields are largest above the antenna itself with attenuation along the propagating axis; however, the fields do not decay to zero. Indeed, the field at the antenna termination is only 20 dB below the peak and there is a non-zero field off the antenna as expected with such a simple antenna configuration. Since the developed antenna field is not fully decayed at the termination, a small standing wave is established (note the ripples in the  FIG. 14  and  FIG. 15  near-zone fields) and this consequently causes gain fluctuations as a function of frequency. Note further that the present invention antenna near-zone fields are very similar to those of the Menzel antenna. 
     FIG. 17  in the drawings shows a polar radiation pattern diagram for a  FIG. 5   a  version of the present invention antenna, a version having the shorting conductors along path  503  spaced at 1.5 millimeter intervals during 6.7 gigahertz operation.  FIG. 18  shows a similar diagram for a  FIG. 5   b  antenna. When the  FIG. 17  and  FIG. 18  drawings are compared, and the scales are adjusted to be the same, it becomes apparent that the  FIG. 5   b  antenna has the same beam width as that for the  FIG. 5   a  antenna. This indicates the rate of leakage with the metal region  503  in  FIG. 5   a  removed is approximately the same as that when this region is present. 
     FIG. 19  in the drawings shows the combination of two present invention antennas in a curved antenna array. This embodiment of the invention illustrates the fact that straight line arrangements of the antenna are not a requirement of the invention, that antenna cooperation is feasible in a relatively small overall space. The inside radius and outside radius coupling of feeder transmission line elements to the  FIG. 19  antennas is worthy of note in the  FIG. 19  drawing. A typical radiation pattern for the  FIG. 19  antenna is shown in the  FIG. 20  drawing. Notably the reduced physical size arrangement of the present invention appears to reduce the degree of mutual coupling between antennas in an array such as that shown in  FIG. 19  and in larger arrays. 
   While the apparatus and method herein described constitute a preferred embodiment of the invention, it is to be understood that the invention is not limited to this precise form of apparatus or method and that changes may be made therein without departing from the scope of the invention, which is defined in the appended claims.