Abstract:
A method and beam-forming network for generating null-filled antenna patterns that approximate double-sided co-secant-squared antenna patterns is disclosed. The method comprises the steps of coupling an input signal to a plurality of output ports arranged in an array through at least one coupler element wherein an amplitude distribution at the output ports follows substantially a ramp function. In another aspect of the invention, electrical phase of the output signals are phase adjusted such that the output phase values are substantially the same.

Description:
BACKGROUND OF THE INVENTION  
         [0001]    The present invention relates to antenna technology and more specifically to methods and beam-forming networks for approximating broadband co-secant squared (CSC 2 ) antenna patterns.  
           [0002]    Cosecant-squared (CSC 2 ) antenna patterns were initially developed for radio guidance systems used to detect and control approaching aircraft. CSC 2  patterns are advantageous as an approaching plane flying at a constant altitude will receive a constant signal strength from the transmitting antenna. CSC 2  antennas are also useful in cellular communication systems for the same reason. In cellular communications, a plurality of antennas are mounted on towers such that each antenna provides transmission and reception of signals within a designated spatial coverage area. Conventionally, three antennas, each covering a sector of 120 degrees, are sufficient for providing acceptable coverage for users within a five-mile radius of a cellular tower. In this case, each sector antenna, with an appropriate down-tilt to account for the elevated height of the antenna, provides constant signal strength for the users on the ground.  
           [0003]    In newer point-to-multipoint systems or local multipoint distributed systems (LMDS), antennas with CSC 2  patterns are also desirable. However, in such systems the antenna pattern is typically non-symmetrical, i.e., single-sided, as there is little need for signal detection above the height of the antenna. However, single-sided CSC 2  antenna exhibit undefined antenna characteristics above the horizontal plane and creates unexpected signal detection responses. Thus, double-sided CSC 2  patterns are more desirable, particularly when the antenna height is on a building or tower that is lower than the customer or user sites.  
           [0004]    Both single- and double-sided CSC 2  pattern sector antennas for cellular systems are conventionally generated by feeding a network of waveguide elements with a waveguide feed at millimeter wave frequencies. Although these networks do not generate a CSC 2  pattern they do generate a “null-filled” pattern that approximates a desired CSC 2  pattern. However, these networks are difficult to construct as precise amplitude and phase control of the waveguide outputs is necessary. Thus, the conventional beam-forming networks are limited to a narrow band of frequencies as it is difficult to implement precise amplitude and phase control in each waveguide over a broad range of frequencies.  
           [0005]    Hence, there is a need for a method for creating beam-forming networks that produce an approximation of a CSC 2  antenna pattern that is easy to implement and maintains the desired CSC antenna characteristics over a broad frequency range. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0006]    In the drawings:  
         [0007]    [0007]FIGS. 1 a  and  1   b  illustrate conventional beam-forming networks;  
         [0008]    [0008]FIG. 1 c  illustrates a graph of a desired near-field amplitude/phase characteristics of a far-field double-sided CSC 2  antenna pattern;  
         [0009]    [0009]FIG. 2 a  illustrates a graph of a near-field amplitude pattern for a summing network;  
         [0010]    [0010]FIG. 2 b  illustrates a graph of a near-field amplitude pattern for a low-gain difference network;  
         [0011]    [0011]FIG. 2 c  illustrates a graph of a combined sum and difference network near-field amplitude pattern;  
         [0012]    [0012]FIG. 2 d  illustrates a first exemplary embodiment of a beam-forming network in accordance with the principles of the invention;  
         [0013]    [0013]FIG. 3 a  illustrates a second exemplary embodiment of a beam-forming network in accordance with the principles of the invention;  
         [0014]    [0014]FIG. 3 b  illustrates a graph of an amplitude distribution pattern for the beam-forming network illustrated in FIG. 3 a;    
         [0015]    [0015]FIG. 3 c  illustrates a far-field antenna pattern produced using the exemplary embodiment shown in FIG. 3 a;    
         [0016]    [0016]FIG. 4 a  illustrates a second aspect of the beam-forming network shown in FIG. 3 a;    
         [0017]    [0017]FIG. 4 b  illustrates a another aspect of the beam-forming network shown in FIG. 3 a;    
         [0018]    [0018]FIG. 5 a  illustrates a third exemplary embodiment of a beam forming network in accordance with the principles of the invention;  
         [0019]    [0019]FIG. 5 b  illustrates a graph of an amplitude distribution pattern for the beam-forming network illustrated in FIG. 5 a ; and  
         [0020]    [0020]FIG. 6 illustrates a still another near-field amplitude distribution in accordance with the principles of the invention. 
     
    
       [0021]    It is to be understood that these drawings are solely for purposes of illustrating the concepts of the invention and are not intended as a definition of the limits of the invention. The embodiments shown in FIGS. 1 through 6 and described in the accompanying detailed description are to be used as illustrative embodiments and should not be construed as the only manner of practicing the invention. It is to be understood that these drawings are for purposes of illustrating the concepts of the invention and are not to scale. Also, the same reference numerals, possibly supplemented with reference characters where appropriate, have been used to identify similar elements.  
       DETAILED DESCRIPTION OF THE INVENTION  
       [0022]    [0022]FIG. 1 a  illustrates a conventional beam-forming network  100 . In this network, a signal applied to input feed  110  is coupled to each of the output ports by dividing the input signal using couplers to route the signal from the input to the outputs. In this illustrative example, a signal applied to input feed  110  is divided using power divider, coupler or splitter,  111 , and coupled into branches  112 ,  113 , respectively. The divided signal in each of branches  112 ,  113  is further divided, using dividers, into branches  114 ,  116 ,  115  and  117 , respectively. The signal in branches  114 ,  116 ,  115  and  117  is next divided by power dividers to form branches  118 ,  120 ,  122 ,  124 ,  119 ,  121 ,  123 , and  125 . Branches  118 ,  120 ,  122 ,  124 ,  119 ,  121 ,  123 , and  125  then provide the signal contained in each branch to a radiating elements, such as horns, waveguides, slots, dipoles, etc., (not shown). In this conventional network, substantially the same amplitude is present at the outputs of branch element  118 ,  120 ,  122 ,  124 ,  119 ,  121 ,  123 , and  125 .  
         [0023]    [0023]FIG. 1 b  illustrates a second conventional beam-forming network  150 . In this network, a signal applied to input feed  110  is divided, using a multiple-output divider, into branches  162  through  176 . In this second conventional network substantially the same amplitude and phase is present at each output port.  
         [0024]    [0024]FIG. 1 c  illustrates a desired conventional near-field amplitude and phase distribution  180  for the conventional beam-forming networks shown in FIGS. 1 a  and  1   b . In this case, each output port has the same amplitude output, represented by graph  185 , and substantially the same phase, represented as graph  190 . The desired near-field amplitude distribution and phase distribution produces a “null-filled” antenna pattern that approximates a CSC 2  antenna pattern. As will be understood, mutual coupling of the output amplitudes is not shown in order to illustrate the desired amplitude distribution. In fact, significant effort is required to reduce the mutual coupling that occurs.  
         [0025]    To achieve the desired the amplitude and phase distributions the beam-forming networks shown must be finely detailed to provide substantially constant signal loss and similar phase change in each signal path. Hence, as frequency of operation changes, the amplitude and electrical phase distribution must be re-determined and readjusted. Thus, for off-frequency or out-of-band operation, the far field antenna characteristics vary significantly as the near-field amplitude and electrical phase distribution characteristics change. Thus, the antenna patterns formed by conventional networks shown are typically limited frequency of operation.  
         [0026]    [0026]FIG. 2 a  illustrates the amplitude distribution  200  of a conventional network that sums the outputs of either network shown in FIG. 1 a  or  1   b . In this case, each element of the beam-forming network provides a summing network with a substantially equal amplitude value. In this case, the amplitude distribution of the summing network may be a constant value. The tapered signal  200  shown in FIG. 2 a  may be obtained by adjusting the signal amplitudes at the ends of the array.  
         [0027]    [0027]FIG. 2 b  illustrates the amplitude distribution  225  of a low-gain difference network. The difference network amplitude distribution may be obtained by altering the phase, by 180 degrees, of one-half of the network elements shown in either FIG. 1 a  or  1   c.    
         [0028]    [0028]FIG. 2 c  illustrates an amplitude distribution  230  of a network that combines, mathematically, the amplitude distributions shown in FIGS. 2 a  and  2   b . In this case, the amplitude distribution produces a ramp-like amplitude distribution  235  and a smaller distribution  240  in the shape of a bump. As will be understood, the far-field antenna patterns of the combined summing and difference network combine to produce a “null-filled” antenna pattern that approximates a CSC 2  pattern.  
         [0029]    [0029]FIG. 2 d  illustrates a first exemplary embodiment  250  of the invention for producing the near-field amplitude distribution pattern shown in FIG. 2 c  in accordance with the principles of the invention. In this embodiment, an input signal applied to input port  110  is divided or split by divider  255  such that a known portion is provided to a first network  260  and the remainder is provided to a second network  280 . The signal applied to network  260  is subsequently divided using dividers  261 - 268  such that the amplitude at each port is progressively less. In preferred embodiment, divider  110  and subsequent dividers  261 - 268  are 3 dB dividers, which divide the signal applied to an input port between two output ports. In this preferred embodiment the output of network  260  is a ramp-like function having one-eighth (⅛ th ) energy of the signal applied at input port  110  at output ports  279 - 279 . The output then is decreased such that one-sixteenth ({fraction (1/16)} th ), one-thirty-second ({fraction (1/32)} nd ), one sixty-fourth ({fraction (1/64)} th ) and one-one hundred twenty-eight ({fraction (1/128)} th ) of the energy of the signal applied at input port  110 , is present at ports  276 ,  275 ,  274  and  273  respectively. And one-two-hundred fifty-sixth ({fraction (1/256)} th ) of the energy of the signal applied at input port  110  is present at output ports  271  and  272 .  
         [0030]    With regard to second network  280 , the applied signal is divided using dividers  281 - 294 , such that the amplitude distribution has one portion that exhibits a ramp-like function and one portion that is substantially symmetrical. In a preferred embodiment, dividers  281 - 294  are 3 dB dividers or splitters. In this preferred embodiment, the signal present at ports  295  and  296  is one-eight (⅛ th ) and one-quarter (¼ th ), respectively, the signal energy applied at input port  110 . As will be understood, divider  283  is used as a combiner wherein the signals applied to two output ports are combined as a single output, which is applied to port  296 .  
         [0031]    With regard to signal applied to divider  284 , this signal is divided such that the amplitude distribution at the remaining ports, represented as  299   a - 299   k  is substantially symmetrical. For example, using 3 dB splitters for dividers  284 - 294 , the amplitude distribution of this portion of network  280  is such that of the signal energy applied to port  110 , one-thirty-second is present at port  299   e , one sixth-fourth ({fraction (1/64)} th ) is present at ports  299   d ,  299   f - 299   h , one one-hundred twenty eight ({fraction (1/128)} th ) is present at ports  299   c ,  299   i  and one-two hundred fifty-sixth ({fraction (1/256)} th ) is present ports  299   a ,  299   b ,  299   j , and  299   k . As will be understood, the length of each of the branches shown is substantially the same in order to provide similar signal loss in each branch  
         [0032]    [0032]FIG. 3 a  illustrates an embodiment of a nine-element beam-forming network in accordance with the principles of the present invention that produces a near-field ramp-like amplitude function that approximates the amplitude distribution shown in FIG. 2 c . In this embodiment of the invention, a signal applied to input waveguide feed  110  is selected divided, using power dividers, couplers or splitters, to provide a known signal power at each output port. In this case, a signal applied to input port  110  is divided, by divider  310 , such that a known portion of the input signal energy is applied directly to output  320  and the remaining portion is applied to divider  322 . The signal energy applied to divider  332  is further divided such that a known portion of the signal energy is applied to divider  324  and the remaining portion is applied to divider  332 . The signal energy applied to divider  324  is then divided and applied to outputs  325  and  330 . The signal energy applied to divider  332  is further divided such that a known portion of the signal energy is applied to divider  334  and the remaining energy is applied to divider  342 . The signal energy applied to divider  334  is further divided such that a known portion is applied to output port  335  and the remaining applied to output port  340 . Similarly, the energy applied to divider  342  is further divided such that a known portion is applied to divider  344  and the remaining energy applied to divider  346 . The signal energy applied to divider  344  is further divided such that a known portion of the energy is applied to output port  345  and the remaining energy is applied to output port  350 . This signal energy applied to divider  346  is divided such that a known portion of the signal energy is applied to output port  355  and the remaining energy is applied to output port  360 .  
         [0033]    Hence, in one aspect of the invention, progressively less of the applied signal energy is applied to each output port. Accordingly, dividers  310 ,  322 ,  332 ,  324 ,  332 ,  334 ,  344  and  336  may be individually adjusted such at any combination of signal energies may be the output from a corresponding port.  
         [0034]    In a preferred embodiment of the invention, dividers  310 ,  322 ,  324 ,  332 ,  334 ,  342 ,  344 , and  346  are elements that divide the input signal energy or power substantially in half. For example, divider elements may be 3 dB splitters that provide one-half the input signal energy or power to each of two output ports. In the preferred embodiment of the invention shown in FIG. 3 a , one-half (½) the input signal energy is output at port  320 , one-eighth (⅛ th ) the input signal energy is output at each of ports  325  and  330 , one-sixteenth ({fraction (1/16)} th ) the input signal energy is output at each of ports  335  and  340  and one-thirty-second ({fraction (1/32)} nd ) the input signal energy is output at each of ports  345 ,  350 ,  355  and  360 . Accordingly, the amplitude distribution having a ramp-like characteristic or function that approximates the amplitude distribution of a network employing a sum and difference network is obtained.  
         [0035]    [0035]FIG. 3 b  illustrates an amplitude distribution  370  of the preferred embodiment of the beam-forming network shown in FIG. 3 a . In this case, the amplitude distribution follows a quasi-exponential ramp-like function, represented by line  375 . In this case, the amplitude distribution exhibits a flattened lower end represented by the quantized or digitized values of amplitude at output ports  345 ,  350 ,  355 , and  360 .  
         [0036]    [0036]FIG. 3 c  illustrates the “null-filled” far-field antenna pattern  385  generated by the preferred embodiment of the network  300  shown in FIG. 3 a  having an amplitude distribution shown in FIG. 3 b  compared to an ideal CSC 2  pattern  390 .  
         [0037]    As will be understood in the art, network  300  is fabricated such that the physical length of each branch between input port  110  and output ports  320 - 360  is substantially equal. In this manner, the signal path, and associated losses, are substantially the same. Furthermore, it is desirable that the electrical phase at each output port is also substantially the same.  
         [0038]    [0038]FIG. 4 a  illustrates a second aspect  400  of the present invention. In this aspect, phase compensators  410 ,  415 ,  420 ,  435 ,  430  and  435  are appropriately located in each waveguide branch of the beam-forming network shown in FIG. 1 a . In this aspect, phase compensators  410 ,  415 ,  420 ,  435 ,  430  and  435  may be used to compensate for anomalies in the output electrical phase of each waveguide branch that may be introduced by imperfections or inaccuracies in the fabrication of network waveguide elements.  
         [0039]    [0039]FIG. 4 b  illustrates a third aspect  450  of the present invention. In this aspect, phase compensators may be introduced at each output port. In this aspect, the phase compensators  455 ,  460 ,  465 ,  470 ,  475 ,  480 ,  485 ,  490 , and  495  may adjust the phase to provide substantially equal electrical phase at each output port. Although not shown, it will be understood that phase compensators,  410 - 435 , shown in FIG. 4 a  or FIG. 4 b , may be mechanically or electrically adjusted to provide substantially the same electrical phase at each output.  
         [0040]    [0040]FIG. 5 a  illustrates another embodiment  500  of a 9-element beam-forming network in accordance with the principles of present invention. In this embodiment, a signal applied to input port  110  is divided by divider  310 , such that a known portion of the applied signal energy or power is applied to output port  320  and the remainder is applied to divider  510 . The signal applied to divider  510  is then divided such that the known portion of signal energy is applied to output  325  and the remainder is applied to divider  515 . The remaining signal energy is then successive applied to dividers  515 ,  520 ,  525 ,  530 ,  535 , and  540 , such that progressively less amounts of the applied signal are output at ports  325 ,  330 ,  335 ,  340 ,  350 ,  355 , and  360 , respectively.  
         [0041]    In another preferred embodiment of the invention, dividers  310 ,  510 ,  515 ,  520 ,  525 ,  530 , and  535  are 3 db dividers or splitters that distribute one-half the signal energy or power applied to the input port to each of the two output ports. In this case, one-half (½) the input signal energy is output on port  320 , one-quarter (¼) the input signal energy is output on port  325 , one-eighth (⅛ th ) the input signal energy is output on port  330 , etc.  
         [0042]    [0042]FIG. 5 b  illustrates an amplitude distribution of the preferred embodiment of the beam-forming network shown in FIG. 5 a . In this case, the amplitude distribution is a monotonically increasing ramp function, represented by line  555 , through the quantized or digitized amplitude values of each output port.  
         [0043]    It will be appreciated that the waveguide length shown in FIG. 5 a  between the input port  110  and each output port is substantially equal. In this case, the signal loss in each waveguide is substantially the same. It is further desirable that the electrical phase at each output port be substantially the same.  
         [0044]    In another aspect of the invention (not shown), and similar to that shown in FIG. 4 a , phase compensators may be incorporated before each divider or coupler or splitter to compensate for phase alterations induced by imperfections or inaccuracies in the fabrication of the network branch elements. Each phase compensator may be individually adjusted, by electrical or mechanical means, to produce electrical phase values at the output ports that are substantially the same. In another aspect (not shown), phase compensators may be implemented at each output port.  
         [0045]    In another aspect of the invention, a near field amplitude pattern generated by the network shown in either FIG. 3 a  or FIG. 5 a  may be a linear ramp function, i.e., triangular or sawtooth pattern. In this case, the amplitude value at each output port may be determined, for example, as:  
         A   i     =       A   1     -           A   1     -     A   l       n     *   i                   (     i   =     2                 …                 n       )                               
 
         [0046]    where A i  is the amplitude at the i th  antenna port;  
         [0047]    A 1  is the greatest amplitude output of the beam-forming network  
         [0048]    A l  is the lowest output amplitude of the beam-forming network; and  
         [0049]    n is the number of elements in the array.  
         [0050]    [0050]FIG. 6 illustrates an amplitude distribution  600  in accordance with the principles of the invention. In this case, the amplitude values at each output port  320 - 360  follows a ramp or triangular or sawtooth function represented by line  610 .  
         [0051]    As will be understood in the art, the amplitude values referred to herein may be represented as measures of signal power. Accordingly, calculations for determining desired output signal values may be performed using the well-known mathematics of logarithms.  
         [0052]    Fundamental novel features of the present invention have has been shown, described, and pointed out as applied to preferred embodiments. It will be understood that various omissions and substitutions and changes in the apparatus described, in the form and details of the devices disclosed, and in their operation, may be made by those skilled in the art without departing from the spirit of the present invention. For example, although the present invention has been described with regard to antenna elevation patterns, it would be understood that horizontal or azimuth antenna patterns may similarly be developed in accordance with the principles of the invention.  
         [0053]    It is also expressly intended that all combinations of those elements that perform substantially the same function in substantially the same way to achieve the same result are within the scope of the invention. Substitutions of elements from one described embodiment to another are also fully intended and contemplated.