Abstract:
A digital receiver (e.g., an ATV or HDTV receiver) which receives digital signals (e.g., ATV or HDTV signals), and which includes an equalizer for equalizing multipath channels having known co-channel interference (e.g., co-channel NTSC interference) present therein. A co-channel interference rejection filter is inserted in the digital receiver upstream of the equalizer, and the equalizer is modified in such a manner that it will not attempt to equalize the co-channel interference. Thus, co-channel interference cancellation is done primarily by the rejection filter, while multipath equalization (correction) is the exclusive function of the equalizer. This leads to better co-channel performance than that which can be obtained by relying on the equalizer alone to perform both functions. The co-channel interference rejection filter is preferably a multi-tap filter having fixed filter coefficients which are designed to optimize cancellation of the known co-channel interference.

Description:
BACKGROUND OF THE INVENTION 
     The present invention relates generally to digital receivers which operate in an environment in which known co-channel interference is present, and, more particularly, to a digital television receiver, such as an HDTV receiver, which utilizes a rejection filter for cancellation of known co-channel interference, e.g., co-channel NTSC interference, and an equalizer for equalizing multipath channels without attempting to equalize the known co-channel interference. 
     The Federal Communications Commission (FCC) has recently approved an advanced television (ATV) standard which encompasses high definition television (HDTV) and standard definition television (SDTV) signals for terrestrial broadcasting. The HDTV signals will be encoded in accordance with the MPEG-2 coding protocol (as described in the ISO/IEC 13818 document), where “MPEG” is an acronyn for the “Moving Pictures Experts Group” which proposed this coding standard. The RF transmission scheme which will likely be used is the trellis-coded 8-VSB (Vestigial SideBand) system developed by Grand Alliance member Zenith Electronics. This system is described in detail in a publication entitled “VSB Transmission System: Technical Details”, Feb. 18, 1994, the disclosure of which is incorporated herein by reference. 
     The FCC will require that ATV signals initially (for at least several years) be broadcast using currently unused analog NTSC television channels (sometimes referred to as “taboo” channels), since ATV broadcasting systems will, at least for this initial period, have to co-exist with conventional analog NTSC broadcasting systems. The resultant simultaneous broadcasting of digital ATV and analog NTSC television signals is oftentimes referred to as “simulcasting”. A practical HDTV receiver must be capable of cancelling the resultant co-channel NTSC interference without excessively enhancing noise, in order to function properly. In this regard, a number of different solutions have been previously proposed, as summarized below. 
     An 8-VSB system developed by Zenith Electronics combats co-channel NTSC interference by using a comb filter in the HDTV receiver to introduce nulls in the digital spectrum at the frequency locations of the NTSC picture, color, and sound carriers. When co-channel NTSC interference is present at the HDTV receiver, the comb filter is treated as a partial response channel in cascade with the trellis decoder. A significant drawback of this approach to combating co-channel NTSC interference is that the performance of the comb filter, and thus, the overall performance of the HDTV receiver, is significantly degraded when co-channel NTSC interference and a high level of additive white Gaussian noise (AWGN) are present in the received signal. This is because the AWGN does not remain white after it is filtered by the comb filter, but gets “colored”, meaning that the noise samples are not taken independently of each other. This, in turn, adversely affects the performance of the trellis decoder, which is optimized for performance in an AWGN channel. Another significant drawback of this approach to combating co-channel NTSC interference is that the comb filter must be switched out in the absence of co-channel NTSC interference, because it would otherwise excessively enhance noise. 
     U.S. Pat. No. 5,291,291, issued to Eilers, discloses an ATV system with reduced co-channel NTSC interference, in which the NTSC receiver subjective random noise sensitivity characteristic is utilized to shape the ATV transmitter power curve. A complementary filter is incorporated in the ATV receiver for compensating for the shaped ATV transmitter power curve. The shaped ATV power curve emphasizes signals at the frequencies where the NTSC subjective random noise sensitivity is low and deemphasizes signals at the frequencies where NTSC subjective sensitivity is high. Significant drawbacks of this approach to combating co-channel NTSC interference are that it requires modification of the ATV transmission system, and requires costly and complex modifications of the ATV receiver. 
     U.S. Pat. Nos. 5,452,015 and 5,512,957, both of which issued to Hulyalkar, and both of which are assigned to the assignee of the present invention, disclose an ATV system including an encoding/transmission system which includes a “bi-rate” control block (to select between 8-VSB and 4-VSB modulation) and respective 8-VSB and 4-VSB “trellis-precoding” blocks, and an ATV receiver having a decoder which is designed to process the subset-limited trellis pre-coded ATV signal which is transmitted by the transmitter. The disclosures of these two patents are incorporated herein by reference. 
     The Hulyalkar ATV receiver utilizes a co-channel NTSC inteference rejection filter and a decoder which processes co-channel NTSC interference in such a manner as to produce a residual interference spectrum which is as flat as possible at the output of the rejection filter. The filter exploits the fact that only the picture and the sound carriers need to be sufficiently attenuated and cancels co-channel NTSC interference with only a small degradation in performance when AWGN is present. Drawbacks of this approach to combating co-channel NTSC interference are that it requires the use of a “subset-limited trellis-coding” precoder in the transmitter of the ATV transmission system (thus changing the transmission stream) and a corresponding decoder and rejection filter in the receiver of the ATV transmission system. 
     U.S. Pat. No. 5,572,249, issued on Nov. 5, 1996, the disclosure of which is incorporated herein by reference, the inventor of which is the present inventor, and the assignee of which is the assignee of the present invention, discloses a filter which can be used for co-channel NTSC inteference cancellation in an ATV system, without introducing excessive noise enhancement. More particularly, this filter could be used in a precoder in the ATV transmitter and the same filter used as a co-channel NTSC interference rejection filter in the ATV receiver. Of course, this entails the same drawbacks discussed above in connection with the Hulyalkar patents. 
     Based on the above and foregoing, it can be appreciated that there presently exists a need in the art for a digital receiver which overcomes the above-described drawbacks of the presently available technology. More particularly, there presently exists a need in the art for a digital receiver, e.g., an HDTV or ATV receiver, which is provided with a co-channel interference rejection filter which enables the realization of an ATV system in which the rejection filter in the receiver can be used alone to cancel co-channel interference, without requiring a precoder in the transmitter and without requiring any change in the transmitted bitstream. The present invention fulfills this need in the art. 
     SUMMARY OF THE INVENTION 
     The present invention encompasses a digital receiver (e.g., an ATV or HDTV receiver) which receives digital signals (e.g., ATV or HDTV signals), and which includes an equalizer for equalizing multipath channels having known co-channel interference (e.g., co-channel NTSC interference) present therein. In general, although the primary function of the equalizer is to equalize the multipath channels, in the presence of co-channel interference, the equalizer will normally attempt to cancel the co-channel interference as well. However, in accordance with one aspect of the present invention, a co-channel interference rejection filter is inserted in the digital receiver upstream of the equalizer, and the equalizer is modified in such a manner that it will not attempt to equalize the co-channel interference. Thus, co-channel interference cancellation is done primarily by the rejection filter, while multipath equalization (correction) is the exclusive function of the equalizer. This leads to better co-channel performance than that which can be obtained by relying on the equalizer alone to perform both functions. The co-channel interference rejection filter is preferably a multi-tap filter having fixed filter coefficients which are designed to optimize cancellation of the known co-channel interference. A suitable such rejection filter which can be employed in the practice of the present invention is the one disclosed in the above-referenced U.S. Pat. No. 5,572,249. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     These and various other features and aspects of the present invention will be readily understood with reference to the following detailed description read in conjunction with the accompanying drawings, in which: 
     FIG. 1 is a block diagram of a conventional decision feedback equalizer; 
     FIG. 2 is a block diagram of a combination which includes a co-channel interference rejection filter and a modified decision feedback equalizer which constitutes a first preferred embodiment of the present invention; and, 
     FIG. 3 is a block diagram of a combination which includes a co-channel interference rejection filter and a modified decision feedback equalizer which constitutes a second preferred embodiment of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     With reference now to FIG. 1, a conventional decision feedback equalizer (DFE)  20  which is utilized to correct multipath and co-channel interference in conventional digital television receivers will now be described. The transmitted data stream, denoted a k , is an 8-VSB signal, which has one of eight different discrete levels, i.e., −7, −5, −3, −1, +1, +3, +5, and +7, where the subscript k represents the time index of the signal sampled at the A/D sampling rate. The input to the equalizer  20 , i.e., the received data stream, denoted r k , is the convolution of the transmitted data stream a k  with the unknown multipath channel h k  plus the additive noise, denoted n k , and the co-channel interference, denoted i k , as defined by the following equations (1) and (2): 
     
       
           r   k   =h   k   *a   k   +n   k   +i   k   (1) 
       
     
     
       
         
           
             
               
                 
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                           - 
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                        
                       
                         
                           h 
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                          
                         
                           a 
                           
                             k 
                             - 
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                       i 
                       k 
                     
                   
                 
               
               
                 
                   ( 
                   2 
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     where * denotes convolution. Since the equalizer  20  functions to restore the transmitted data stream a k , it will attempt to equalize the multipath channel component h k  as well as the co-channel interference component i k . 
     More particularly, the equalizer  20  includes a forward filter  22  which is a finite impulse response (FIR) filter having a plurality Lf of taps and respective filter coefficients f 0 -f Lf−1 , where the delay between each tap is preferably equal to one symbol interval of the transmitted data stream, which is the reciprocal of the A/D sampling rate of the transmitted data stream, which in the case of the 8-VSB signal is 10.76 MHz. The equalizer  20  also includes a feedback filter  24  which is an FIR filter having a plurality Lb of taps and respective filter coefficients b 1 -b Lb , where the delay between each tap is preferably equal to one symbol interval of the transmitted data stream, which is the reciprocal of the A/D sampling rate of 10.76 MHz. The equalizer  20  further includes an adder (or subtractor)  26  which subtracts the output of the feedback filter  24  from the output of the forward filter  22 . The output of the forward filter is f k *r k , and the output of the feedback filter  24  is b k *{circumflex over (a)} k . The output of the adder  26  is {tilde over (a)} k =f k *r k −b k *{circumflex over (a)} k . The output {tilde over (a)} k  of the adder  26  is taken as the output of the equalizer  20 , and is supplied to the trellis decoder (not shown) of the digital receiver. The equalizer  20  also includes a slicer  28  which “slices” the output {tilde over (a)} k  to one of the eight possible discrete values of the 8-VSB signal, i.e., −7, −5, −3, −1, +1, +3, +5, and +7. 
     With reference now to FIG. 2, a preferred embodiment of the present invention will now be described. More particularly, in accordance with this embodiment of the present invention, a digital receiver includes a co-channel interference rejection filter  30  which functions to cancel known co-channel interference in a received data stream r k , and a decision feedback equalizer (DFE)  32  which functions to equalize multipath channels of the data stream. In accordance with an aspect of the present invention, the equalizer  32  is designed so that it will not attempt to equalize the co-channel interference. In a presently contemplated implementation of this embodiment of the present invention, the digital receiver is an HDTV receiver, the known co-channel interference is co-channel NTSC interference, and the received data stream r k  is the convolution of the transmitted data stream a k  with the unknown multipath channel h k  plus the additive noise, denoted n k , and the co-channel interference, denoted i k , as defined by the equations (1) and (2) above, where the transmitted data stream a k  is an 8-VSB signal. However, this is not limiting to the present invention, as the present invention has applicability to any digital receiver which receives signals corrupted with known co-channel interference. 
     With continuing reference to FIG. 2, the received data stream r k  is input to the rejection filter  30 . In the most general case, the rejection filter  30  is a finite impulse response (FIR) filter having a plurality Lg of taps and respective filter coefficients g 0 -g Lg−1 . The filter coefficients g 0 -g Lg−1  can be selected in any manner which results in cancellation (or reduction) of the known co-channel interference, without excessively enhancing the noise. For example, the co-channel inteference rejection filter disclosed in the above-referenced U.S. Pat. No. 5,572,249 can suitably be employed, in which the filter coefficient g 0  is selected to have a value of 1 and all the other filter coefficients selected to have values less than 1, i.e., the filter is causal. For purposes of the present disclosure, it will be assumed that the rejection filter  30  (or g k ) is a co-channel interference rejection filter of this type. However, it should be clearly understood that this is not limiting to the present invention. 
     The output of the rejection filter  30 , denoted y k , (which is the input to the DFE  32 ), is defined by the following equation (3): 
     
       
           y   k   =c   k   *h   k   +i   k   *g   k   +n   k   *g   k ,  (3) 
       
     
     where * denotes convolution, c k  denotes the response of the rejection filter  30  to the transmitted data stream a k , i k  denotes the known co-channel interference, n k  denotes the additive noise present in the received signal, h k  denotes the unknown multipath channel, and the subscript k denotes the time index of the received signal sampled at the A/D sampling rate, e.g., 10.76 MHz. The rejection filter  30  is designed to minimize the second quantity (i k *g k ) in the above equation (3), i.e., the co-channel interference component, while not excessively enhancing the third quantity (n k *g k ) in the above equation (3), i.e., the noise component. 
     The DFE  32 , in accordance with the present invention, is designed to not attempt to restore the transmitted data stream a k , because in order to do so, the DFE  32  would have to undo the effect of the rejection filter  30 . Instead, the DFE  32  functions to reconstruct the sequence c k , which is defined by the following equation (4):                c   k     =         g   k     ⋆     a   k       =       ∑     i   =   0         L   g     -   1              g   i          a     k   -   i                     (   4   )                                
     With continuing reference to FIG. 2, the DFE  32  of the present invention has the same architecture as that of the conventional DFE  20  depicted in FIG. 1, with the exception that the DFE  32  of the present invention has a modified slicer  35  whose function will be described below. More particularly, the DFE  32  includes a forward filter  42  which is a finite impulse response (FIR) filter having a plurality Lf of taps and respective filter coefficients f 0 -f Lf−1 , where the delay between each tap is preferably equal to one symbol interval of the transmitted data stream, which is the reciprocal of the A/D sampling rate of the transmitted data stream, which in the case of the 8-VSB signal is 10.76 MHz. Thus, the forward filter  42  of the DFE  32  of the present invention is of the same design as the forward filter  22  of the conventional DFE  20 . The DFE  32  also includes a feedback filter  44  which is an FIR filter having a plurality Lb of taps and respective filter coefficients b 1 -b Lb , where the delay between each tap is preferably equal to one symbol interval of the transmitted data stream, which is the reciprocal of the A/D sampling rate of 10.76 MHz. Thus, the feedback filter  44  of the DFE  32  of the present invention is of the same design as the feedback filter  24  of the conventional DFE  20 . The DFE  32  further includes an adder (or subtractor)  46  which subtracts the output of the feedback filter  44  from the output of the forward filter  42 . Thus, the adder  46  of the DFE  32  of the present invention is of the same design as the adder  26  of the conventional DFE  20 . 
     However, since the input to the forward filter  42  is y k , the output of the forward filter  42  is f k *y k , and since the input to the feedback filter  44  is {circumflex over (c)} k , the output of the feedback filter  44  is b k *{circumflex over (c)} k . Thus, the output of the adder  46 , denoted {tilde over (c)} k , is defined by the following equation (5): 
     
       
           {tilde over (c)}   k   =f   k   *y   k   −b   k   *{circumflex over (c)}   k .  (5) 
       
     
     As will become apparent hereinafter, {circumflex over (c)} k  is the “sliced” version of {tilde over (c)} k . The output {tilde over (a)} k  of the adder  52  is taken as the output of the DFE  32 , and is supplied to the trellis decoder (not shown) of the digital receiver. 
     The DFE  32  operates in the following manner. More particularly, a known periodic sequence, referred to as the “training sequence”, is inserted in the transmitted data stream a k  at the transmitter, in accordance with the MPEG-2 Grand Alliance ATV/HDTV Transmitter Standard, in order to enable the equalizer in the receiver to converge thereon and thereby be synchronized with the subsequent actual data in the transmitted data stream. During this “training sequence”, since a k  is known, and since the rejection filter coefficients g k  are also known, c k  can be easily calculated. However, after the equalizer has converged on the “training sequence”, since a k  is no longer known, then c k  also becomes unknown. The feedback filter  44  still requires the “sliced” version {circumflex over (c)} k  of {tilde over (c)} k  at its input in order to function properly. Thus, since c k  is no longer composed of discrete levels like the transmitted data stream a k , {tilde over (c)} k  can not be sliced in the usual manner. 
     Therefore, without any modification to the standard slicer  28  which is utilized in the conventional DFE  20  (depicted in FIG.  1 ), the “sliced” version {circumflex over (c)} k  of {tilde over (c)} k  can not be supplied the input of the feedback filter  44 , as required. For this reason, the “modified slicer”  35  is included in the DFE  32  of the present invention. The modified slicer  35  includes, in addition to a standard slicer  48 , an additional feedback filter  50  and two additional adders  52  and  54 . The feedback filter  50  is provided with the same filter coefficients g k  as those of the co-channel interference rejection filter  30 . The input to the additional feedback filter  50  is {circumflex over (a)} k , and the output of the feedback filter, denoted d k , is defined by the following equation (6): 
     
       
           d   k   ={tilde over (c)}   k   −{circumflex over (a)}   k .  (6) 
       
     
     The output d k  of the additional feedback filter  50  is applied as the inverted input of the adder  52  and as one of the non-inverted inputs to the adder  54 . Thus, the output {circumflex over (c)} k  of the adder  54  is defined by the following equation (8): 
       {circumflex over (a)}   k   +d   k   ={circumflex over (c)}   k ,  (8) 
     which is the input (as required) to the feedback filter  44 . The output {tilde over (a)} k  of the adder  52  is defined by the following equation (9): 
     
       
           {tilde over (c)}   k   −d   k   ={tilde over (a)}   k ,  (9) 
       
     
     which is taken as the output of the DFE  32  and supplied to the next stage of the receiver, i.e., the trellis decoder. 
     It can be easily seen from equation (4) above that the following relation holds between c k  and a k , assuming g 0  is 1:                a   k     =       c   k     -       ∑     i   =   0         L   g     -   1              g   i          a     k   -   i                     (   10   )                                
     Hence, in the modified slicer  35 , {tilde over (a)} k  can be reconstructed from the equalizer output {tilde over (c)} k  and past decisions {circumflex over (a)} k  as follows:                  a   k     ~     =           c   k     ~     -       ∑     i   =   0         L   g     -   1              g   i            a   ^       k   -   i             =         c   k     ~     -       d   k     .                 (   11   )                                
     {tilde over (a)} k  can be sliced in the normal manner by the standard slicer  48  to give {circumflex over (a)} k . Finally, the output {circumflex over (c)} k  of the adder  54 , which is the input (as required) to the feedback filter  44 , is obtained as follows:                  c   ^     k     =           a   ^     k     +       ∑     i   =   0         L   g     -   1              g   i            a   ^       k   -   i             =         a   ^     k     -       d   k     .                 (   12   )                                
     With reference now to FIG. 3, there can be seen an alternative embodiment of the present invention. The only difference between this embodiment and the one depicted in FIG. 2 is that {tilde over (c)} k  is taken as the output of the DFE  32 , instead of {tilde over (a)} k , and supplied as the input to the trellis decoder. Because the trellis decoder will have {tilde over (c)} k  rather than {tilde over (a)} k  as its input, it will have to be modified. More particularly, with this embodiment, the trellis decoder will have to be implemented as a parallel decision feedback decoder (PDFD), i.e., it will have a separate decision feedback part for each state, such as is disclosed in an article entitled “Delayed decision-feedback sequence estimation”,  IEEE Trans. Commum. , Vol. COM-37, No. 5, pp. 428-436, May 1989, the disclosure of which is incorporated herein by reference. The number of states in the PDFD remain the same as in the original trellis decoder. However, each state has associated with it a best path that is L g −1 symbols long that is used for the metric calculation procedure. The advantage of this embodiment as compared to that of the embodiment depicted in FIG. 2 is that it will suffer less from error propagation and hence will exhibit improved performance. However, the complexity of this implementation is increased due to the additional storage and filtering requirements. 
     Although preferred and alternative embodiments of the present invention have been described in detail hereinabove, it should be clearly understood that many variations and/or modifications of the basic inventive concepts herein taught which may appear to those skilled in the pertinent art will still fall within the spirit and scope of the present invention, as defined in the appended claims. For example, although the present invention has been discussed in the context of simultaneous broadcasting of HDTV/ATV and NTSC television signals wherein co-channel NTSC interference is a concern, it will be readily appreciated that the present invention is equally applicable to the context of HDTV/ATV and SECAM or PAL television signals, or any other conventionally broadcast television signals. Moreover, as was previously noted, the present invention has applicability to any digital receiver which receives signals corrupted with known co-channel interference.