Abstract:
An electronic converter may include transformer with a primary winding and a secondary winding, wherein the primary winding is coupled to an input for receiving a power signal, and wherein the secondary winding is coupled to an output including a positive terminal and a negative terminal for providing a power signal. The converter moreover may include an electronic switch arranged between the input and the primary winding, wherein the electronic switch is configured to control the current flow through the primary winding. Specifically, the converter may include a snubber circuit arranged between the secondary winding and the output.

Description:
FIELD OF THE INVENTION 
       [0001]    The present disclosure relates to electronic converters. 
         [0002]    This disclosure was devised with specific attention paid to the provision of a suppressor or snubber for an electronic converter. 
       DESCRIPTION OF THE RELATED ART 
       [0003]    Electronic converters for lighting sources comprising, for example, at least one LED (Light Emitting Diode) or other solid-state lighting means may provide a direct current output. Such current may be stable or vary in time, for example in order to regulate the light intensity emitted by the lighting source (so called dimming function). 
         [0004]      FIG. 1  shows a possible lighting system comprising an electronic converter  10  and a lighting module  20 , comprising for example at least one LED L. 
         [0005]    Electronic converter  10  usually comprises a control circuit  102  and a power circuit  12  (for example an AC/DC or DC/DC switching supply) and provides as output a direct current through a power output  106 . Such a current may be stable, or else may vary in time. For example, control circuit  102  may set, via a reference channel I ref  of power circuit  12 , the current required by LED module  20 . 
         [0006]    For example, such a reference channel I ref  may be used in order to regulate the intensity of the light emitted by lighting module  20 . Actually, a regulation of the light intensity emitted by LED module  20  may be generally achieved by regulating the average current flowing through lighting module  20 , for example by setting a lower reference current I ref , or by switching on or off power circuit  12  through a Pulse Width Modulation (PWM) signal. 
         [0007]    However, the case wherein module  20  is supplied with a regulated voltage, i.e. wherein converter  12  is a voltage generator, typically requires a current regulator which is connected in series with lighting sources L, in order to limit the current. In this case, the dimming function may be implemented also via such a current regulator, for example:
       a) by selectively switching on or off such a current regulator via a driving signal, e.g. a PWM signal, or   b) in case of an adjustable current regulator, by setting the reference current of such a current regulator.       
 
         [0010]    Generally speaking there are many types of electronic converters, which are divided mainly into isolated and non-isolated converters. For example, non-isolated electronic converters are “buck”, “boost”, “buck-boost”, “Cuk”, “SEPIC” and “ZETA” converters. On the contrary, isolated converters are for example “flyback”, “forward”, “Half-bridge” and “Full-bridge” converters. Such kinds of converters are well known to the skilled in the art. 
         [0011]    For example,  FIG. 2  shows the circuit arrangement of a flyback converter. 
         [0012]    As is well-known, a flyback converter comprises a transformer T with a primary winding T 1  and a secondary winding T 2 , an electronic switch S, such as for example an n-channel MOSFET transistor (Metal-Oxide-Semiconductor Field-Effect Transistor), or a bipolar or IGBT transistor (Insulated-Gate Bipolar Transistor), a diode D 1  and an output capacitor Co. 
         [0013]    Specifically, transformer T may be modelled as an inductance Lm, connected in parallel with primary winding T 1 , which represents the magnetising inductance of transformer T, and an ideal transformer with a given turn ratio 1:n. 
         [0014]    In the presently considered example, converter  12  receives as input, via two input terminals, a voltage V in , and provides as output, via a supply line  106 , a regulated current i out . Those skilled in the art will appreciate that voltage V in  may also be obtained through an input AC current, for example via a diode or a diode bridge rectifier, and optionally a filtering capacitor. 
         [0015]    Specifically, the first input terminal is connected to the first terminal of primary winding T 1  of transformer T and the second input terminal represents a first ground GND 1 . On the contrary, the second terminal of primary winding T 1  of transformer T is connected through switch S to ground GND 1 . Therefore, switch S may be used to selectively activate the current flow through primary winding T 1  of transformer T. 
         [0016]    On the other hand, the first terminal of secondary winding T 2  of transformer T is connected through a diode D 1  to a first output terminal, which represents power output  106 , and the second terminal of secondary winding T 2  of transformer T is connected directly to a second output terminal, which represents a second ground GND 2 , which due to the isolating effect of transformer T is preferably different from ground GND 1  and is therefore denoted with a different ground symbol. 
         [0017]    Finally, an output capacitor Co is connected in parallel with the output, i.e. between terminals  106  and GND 2 . 
         [0018]    Therefore, when switch S is closed, primary winding T 1  of transformer T is connected directly to input voltage V in . This causes an increase of the magnetic flow in transformer T. Therefore, the voltage across secondary winding T 2  is negative, and diode D 1  is inversely biased. In this condition, output capacitor Co provides the energy required by the load, for example by lighting module  20 . 
         [0019]    On the contrary, when switch S is open, the energy stored in transformer T is transferred as flyback current to lighting module  20 . 
         [0020]    As previously mentioned, the control may be in current or voltage. To this purpose, a control unit  112  is typically used which drives switch S so that output voltage V out  or output current i out  is regulated on a desired value, so as for example reference current I ref . To this purpose it is possible to use, as known in itself, a sensor adapted to detect current i out  or voltage V out . 
         [0021]    Typically, control unit  112  drives switch S with Pulse Width Modulation (PWM), wherein switch S is closed during a first operation interval and switch S is opened during a second operation interval. Those skilled in the art will appreciate that such PWM driving and the control of duration of operation intervals are well known, and may be implemented, for example, through a feedback of the output voltage or current through an error amplifier. For example, in the case of a current control, the duration of the first interval is increased until the (average) output current reaches a predetermined threshold. 
         [0022]    Such a PWM driving may involve three different operation modes. Specifically, if the current in the magnetising inductance Lm never reaches zero during a switching cycle, the converter is said to operate in a Continuous Current Mode (CCM). On the contrary, when the current in the magnetising inductance Lm reaches zero during the period, the converter is said to operate in a Discontinuous Current Mode (DCM). Typically, the converter operates in a discontinuous mode when the load absorbs a low current, and in a continuous mode at higher levels of absorbed current. The border between the continuous mode, CCM, and the discontinuous mode, DCM, is reached when the current reaches zero, exactly at the end of the switching cycle. Such a limit case is referred to as Transition Mode (TM). Moreover, there is the possibility of driving the switch with a resonant or quasi-resonant driving, wherein switch S is switched when the voltage across said electronic switch (S) is zero, or when a local minimum is reached. Typically the switching frequency, i.e. the sum of the duration of operation periods, is fixed for a CCM or a DCM driving, and is variable for a quasi-resonant driving. 
         [0023]    However a flyback converter, and generally every switching power supply, comprises parasitic elements. For example, in a flyback converter one of the most influential elements is transformer T, particularly its leakage inductance. For example, in  FIG. 2 , the leakage inductance of transformer T is modelled as an inductance Lr connected in series with the secondary winding T 2  of transformer T. In a similar way, in a forward converter, both the magnetizing inductance Lm and the leakage inductance Lr constitute parasitic elements. Actually such inductances store energy which often cannot be transferred to the load. For example, in a flyback converter the discharge of the energy stored in the parasitic inductance Lr may cause an overvoltage across switch S. Moreover the zeroing of current through switch S cannot take place with zero voltage, which involves switching losses as well. 
         [0024]    Therefore snubbers have been used in the past. Such snubbers are typically divided into the following categories:
       dissipative snubber: a dissipative network comprising passive components, particularly resistors;   non-dissipative passive snubber: a circuit comprising one or several reactive components, for example capacitors, which allow for the recovery of the energy stored in the inductive components; and   non dissipative active snubber: a circuit comprising a passive network and one or several switches.       
 
         [0028]    Snubber circuits also have other advantages, such as:
       Electromagnetic Interference (EMI) is typically reduced; and   the switching of the switch or switches of the switching supply may take place at zero voltage: it is the so-called Zero-Voltage Switching (ZVS).       
 
         [0031]    Details on the operation of passive snubber circuits are described, for example, in P. C. Todd, “ Snubber circuits: Theory, Design and Application ”, Unitrode Corporation, May 1993, the content whereof is incorporated herein by reference. 
         [0032]    Details on the operation of non-dissipative snubber circuits, for example for flyback converters, are described in T. Ninomiya, T. Tanaka, and K. Harada, “ Analysis and optimization of a non - dissipative LC turn - off snubber ,” IEEE Transactions on Power Electronics, vol. 3, no. 2, pp. 147-156, 1988, or in Chih-Sheng Liao, Keyue M. Smedley, “ Design of high efficiency Flyback converter with energy regenerative snubber ”, Conference: Applied Power Electronics Conference and Exposition Annual IEEE Conference—APEC, pp. 796-800, 2008, the contents whereof are incorporated herein by reference. 
         [0033]    Finally, active snubber circuits are described for example in B. Andreycak, “ Active clamp and reset technique enhances forward converter performance ”, Unitrode Power Supply Design Seminar, SEM-1000, pp. 3-1-3-18, 1994 for forward converters, and in Robert Watson, et al., “ Utilization of an Active - Clamp Circuit to Achieve Soft Switching in Flyback converters ”, IEEE Transactions on Power Electronics, V. 11, pp. 162-169, 1996 for flyback converters, the contents whereof are incorporated herein by reference. 
         [0034]    The previously described snubber circuits have common features, as they are located on the primary side of the transformer, and they limit the peak or rise-time of the voltage across the main switch. Therefore, such circuits cannot directly snub effects which are caused by components on the secondary side of the transformer. 
         [0035]    Document EP 1 202 440 A1 discloses a snubber circuit which is located at the secondary side of a transformer. Specifically, the disclosed snubber circuit comprises two diodes, a capacitor and an inductor. According to document EP 1 202 440 A1 the inductor of such a snubber circuit permits that the current of the primary side switch is smoothly and gradually decreased due to the current supplied from the capacitor of the snubber circuit, so that the voltage of the primary side switch increases with a gradient that a voltage ringing is suppressed. 
         [0036]    The inventor has noted that the arrangements disclosed in document EP 1 202 440 A1 have several inconveniences. For example, according to this document, the snubber capacitor on the secondary side is discharged with the same current that will supply the load. Thus, in order to have an important slope reduction of the rise time of the voltage of the primary side switch, a high capacitance is required for the snubber capacitor. However, a high capacitance value implies that also a high inductance is required for the inductor of the snubber circuit in order to charge the snubber capacitor during the switch-on time. 
         [0037]    Moreover, when the output diode starts to conduct, the leakage inductance of the transformer and the primary side switch capacitance will tend to oscillate as is shown e.g. in FIG. 20A of document EP 1 202 440 A1. 
       OBJECT AND SUMMARY 
       [0038]    The object of the invention is to improve snubbing techniques. 
         [0039]    Actually, inventors have observed that it is also possible to provide a snubber comprising components located on the secondary side of the transformer. 
         [0040]    According to the invention, such an object is achieved through an electronic converter having the features set forth in the claims that follow. The claims also concern a related lighting system and method of operating an electronic converter. 
         [0041]    In various embodiments, the electronic converter comprises a transformer with a primary winding and a secondary winding, wherein the primary winding is coupled to an input to receive a power signal, and wherein the secondary winding is coupled to an output comprising a positive terminal and a negative terminal, in order to provide a power signal. 
         [0042]    In various embodiments, the electronic converter moreover comprises an electronic switch located between the input and the primary winding, in order to control the current flow through the primary winding of the transformer. 
         [0043]    In various embodiments, a snubber circuit is located between the secondary winding and the output. 
         [0044]    In various embodiments, a second snubber circuit is associated to the primary winding of the transformer as well, particularly to the electronic switch. Typically, such a snubber circuit comprises a capacitor. However, the operation of the capacitor may also be implemented through the parasitic capacitance of the electronic switch. Therefore, in various embodiments, the sum of the parasitic capacitance of the electronic switch and the capacitance of other possible circuits connected in parallel with the switch is between 10 pF and 1 nF. 
         [0045]    In various embodiments, the snubber circuit on the secondary side of the transformer comprises two diodes connected (directly) in series and a capacitor connected (directly) to the intermediate point between both diodes. For example, in various embodiments the cathode of the first diode is connected (directly) to the positive output terminal, and the anode of the second diode is connected (directly) to the negative output terminal. 
         [0046]    The connection of the capacitor of the snubber circuit on the secondary side of the transformer depends on the kind of converter. Typically, the capacitor of the snubber circuit on the secondary side is connected (directly) to the leakage inductance of the transformer. For example, if the electronic converter is a flyback converter comprising a diode connected between a terminal of the secondary winding and an output terminal, the capacitor of the snubber circuit is connected (directly) to such a terminal of the secondary winding, i.e. the leakage inductance of the transformer. Conversely, the capacitor could be connected (directly) to the output inductor of a forward converter, a Cuk converter or a ZETA converter. 
         [0047]    The previously described snubber circuits allow for the reduction of the voltage across the electronic switch of the converter. For this reason, the converter may be driven in the resonant or quasi-resonant mode, i.e. the electronic converter is closed when the voltage across the electronic switch is zero or has reached a local minimum. 
         [0048]    Specifically, during the switch-off time, the primary side snubber circuit and the secondary side snubber circuit will generate a resonance together with the main leakage inductance of the converter. 
         [0049]    More specifically, when the primary side switch is opened, the first diode of the snubber circuit is used to discharge the capacitor of the snubber circuit and charge the capacitance associated with the primary side switch. Thus, by dimensioning in a convenient manner both capacitance values, the oscillation produced by the switch capacitance and the leakage inductance may be reduced and almost be eliminated. For example, in various embodiments, the capacitance associated with the primary side switch and the capacitor of the snubber circuit on the secondary side have similar values. 
         [0050]    Conversely, when the primary side switch is closed, the second diode of the snubber circuit is used to recharge the capacitor or the snubber circuit on the secondary side. Thus, the capacitance associated with the primary side switch and the capacitor of the snubber circuit on the secondary side should be selected in order to permit that the main leakage inductance of the converter may recharge smoothly the capacitor of the snubber circuit on the secondary side during the switch-on time. For example, such capacitances may be between 100 pF and 1 nF and preferably between 300 pF and 600 pF. 
     
    
     
       BRIEF DESCRIPTION OF THE ANNEXED VIEWS 
         [0051]    The invention will now be described, by way of non-limiting example only, with reference to the enclosed views, wherein: 
           [0052]      FIGS. 1 and 2  have already been described in the foregoing, 
           [0053]      FIGS. 3 to 8  show details of embodiments of snubber circuits according to the present disclosure. 
       
    
    
     DETAILED DESCRIPTION OF EMBODIMENTS 
       [0054]    In the following description, numerous specific details are given to provide a thorough understanding of embodiments. The embodiments may be practiced without one or more of the specific details, or with other methods, components, materials, etc. In other instances, well-known structures, materials, or operations are not shown or described in detail to avoid obscuring aspects of the embodiments. 
         [0055]    Reference throughout this specification to “one embodiment” or “an embodiment” means that a particular feature, structure, or characteristic described in connection with the embodiment is included in at least one embodiment. Thus, the appearances of the phrases “in one embodiment” or “in an embodiment” in various places throughout this specification are not necessarily all referring to the same embodiment. Furthermore, the particular features, structures, or characteristics may be combined in any suitable manner in one or more embodiments. 
         [0056]    The headings provided herein are for convenience only and do not interpret the scope or meaning of the embodiments. 
         [0057]    As previously stated, the present disclosure provides solutions allowing for the implementation of a snubber circuit for a switching supply comprising a transformer, such as for example a flyback, forward, Cuk, SEPIC or ZETA isolated converter. 
         [0058]    In various embodiments, the snubber circuit comprises two sub-circuits. The first circuit is located on the primary side of the transformer, i.e. on the side of the transformer to which one or several switches are coupled to control the current flow in the primary winding of the transformer. On the contrary, the second circuit is located on the secondary side of the transformer, i.e. on the side of the transformer whereto the load is coupled. 
         [0059]      FIG. 3  shows an embodiment of a flyback converter according to the present disclosure. Specifically, the flyback converter shown in  FIG. 3  is substantially based on the converter shown in  FIG. 2 . Therefore, the operation of such a circuit will not be repeated, the attention being focused only on the differences. 
         [0060]    Specifically, as previously stated, the converter comprises two additional circuits, i.e. a first additional circuit  30  on the primary side T 1  of transformer T and a second additional circuit  32  on the secondary side T 2  of transformer T. 
         [0061]    In the presently considered embodiment, the first circuit  30  comprises a capacitance C 1 , which is associated to switch S. Specifically, in the considered embodiment, capacitance C 1  is connected directly in parallel with switch S. Such a capacitance C 1  may be inherent to switch S, i.e. it may be a parasitic capacitance and/or it may be implemented through a capacitor external to switch S. For example, in various embodiments, the sum of the parasitic capacitance of switch S and the capacitance of possible other capacitors connected in parallel with switch S is between 10 pF and 1 nF. 
         [0062]    On the contrary, the second circuit  32  substantially comprises a charge pump. 
         [0063]    Specifically, in the considered embodiment, circuit  32  comprises two diodes D 2  and D 3  connected (directly) in series. In the considered embodiment, such diodes D 2  and D 3  are connected in turn in parallel to the output capacitor Co, i.e. the cathode of diode D 2  is connected (directly) to line  106  and the anode of diode D 3  is connected (directly) to ground GND 2 . Thus, in the embodiment considered, the snubber circuit does not comprise any additional inductive component. 
         [0064]    In the considered embodiment, circuit  32  comprises moreover a capacitor C 2 , which is connected (directly) between the first output terminal of the secondary winding T 2  of transformer T, i.e. to the anode of diode D, and the intermediate point between the diodes D 2  and D 3 , i.e. to the anode of diode D 2  and the cathode of diode D 3 . 
         [0065]    For example, typical values for capacitors are a few hundreds of pF, i.e. the value of capacitances C 1  and C 2  ranges from 100 pF to 1 nF and preferably between 300 pF and 600 pF. Thus, preferably, the capacitances C 1  and C 2  have similar values, and the values should be selected in order to permit that the leakage inductance of the transformer T may recharge smoothly the capacitor C 2  during the switch-on time as will be described in the following. 
         [0066]    In the following a possible embodiment of the driving of such a flyback converter will be described with reference to  FIGS. 4   a  to  4   g  and to  FIGS. 5   a  to  5   g.  Specifically,  FIGS. 4   a  to  4   g  show different equivalent circuit arrangements for different operation periods. On the other hand,  FIGS. 5   a  to  5   g  show typical waveforms respectively for voltage V S  across switch S, voltage V C1  across capacitor C 1 , voltage V C2  across capacitor C 2 , current i C2  flowing through capacitor C 2 , current i D1  flowing through diode D 1 , current i D2  flowing through diode D 2  and current i D3  flowing through diode D 3 . 
         [0067]    Specifically, at a time instant t 0 , switch S is opened. During the following operation period (see  FIG. 4   a ), the energy stored in the magnetising inductance Lm charges capacitor C 1  until its voltage V C1  equals 
         [0000]        V   in   +n ·( V   out   −V   C2 )  (1)
 
         [0068]    If the flyback converter is operated at a Zero Voltage Switching (ZVS), the voltage across capacitor C 1  would be exactly n·V out . 
         [0069]    At a time instant t 1 , the sum of the voltage at secondary winding T 2  and capacitor C 2  is sufficient to switch diode D 2  on. Therefore, during the following operation period (see  FIG. 4   b ), capacitor C 2  stars discharging, while capacitor C 1  keeps on charging. 
         [0070]    At a time instant t 2 , capacitor C 2  is completely discharged. Specifically, the period between instant t 2  and instant t 1  is: 
         [0000]    
       
         
           
             
               
                 
                   
                     
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         [0071]    Therefore, at instant t 2  diode D 2  is opened and diode D 1  is closed. As a consequence, during the following operation period (see  FIG. 4   c ), the converter behaves as a conventional flyback converter, wherein the energy stored in the magnetising inductance Lm is discharged towards output capacitor Co and the load. 
         [0072]    At a time instant t 3 , magnetising inductance Lm is discharged, and as a consequence diode D 1  is opened. Therefore, during the following operation period (see  FIG. 4   d ), capacitor C 1  starts discharging through inductance Lm. This represents a first resonant circuit LC (Lm and C 1 ) which creates an oscillation allowing for ZVS or quasi-ZVS. Specifically, the resonant transition is set by 
         [0000]      √{square root over (Lm·C1)}  (3)
 
         [0073]    At a time instant t 4 , the voltage at secondary winding T 2  is sufficient to switch diode D 3  on. Therefore, during the following operation period (see  FIG. 4   e ), a second oscillation begins as well, being generated by a second resonant circuit LC (Lr and C 2 ). 
         [0074]    Therefore, the sum of the oscillations creates a complex multi-resonant transition which charges capacitor C 2 . 
         [0075]    At a time instant t 5 , switch S is closed. Specifically, in the considered embodiment, this takes place at quasi-ZVS. Therefore, during the following operation period (see  FIG. 4   f ) capacitor C 2  is charged through the remaining energy stored in the leakage inductance Lr. 
         [0076]    At a time instant t 6 , the leakage inductance is discharged and diode D 3  is opened. Therefore, during the following operation period (see  FIG. 4   g ) the converter behaves as a conventional flyback converter, wherein the magnetising inductance Lm is charged. 
         [0077]    Afterwards, the previously described operation periods will be repeated. 
         [0078]    As previously stated, the same snubber circuit may be used in other types of converters. 
         [0079]    Typically, the snubber circuit at the secondary side should be positioned in order to bypass the main rectifying diode, e.g. in case of a single voltage output filter, such as in the Flyback or SEPIC topology, or the filter inductor, e.g. in case of a double current voltage output filter, such as in the Forward, Cuk or Zeta topology. 
         [0080]    For example,  FIG. 6  shows an embodiment of a forward converter. 
         [0081]    Those skilled in the art will appreciate that a forward converter has, on the primary side T 1  of transformer T, the same electrical connection as a flyback converter. On the contrary, on the secondary side, the forward converter comprises an inductor Lo, two diodes D 1  and D 4  and a capacitor Co. Specifically, inductor Lo is connected between the first terminal of the secondary winding T 2  of transformer T and the anode of diode D 1 . The cathode of diode D 1  is connected to line  106 , and diode D 4  is connected between the second terminal of secondary winding T 2  of transformer T, which represents ground GND 2 , and the anode of diode D 1 . Finally, capacitor Co is connected in parallel to the load, i.e. between line  106  and ground GND 2 . 
         [0082]    In this case, circuits  30  and  32  are not modified; what changes is only the connection of circuit  32  on the secondary side T 2  of transformer T. 
         [0083]    Specifically, in the considered embodiment, the cathode of diode D 2  is still connected (directly) to the positive output terminal, i.e. line  106 , the anode of diode D 3  is connected (directly) to ground GND 2  and capacitor C 2  is connected (directly) to the first terminal of the secondary winding T 2  of the transformer, i.e. to the intermediate point between transformer T and inductor Lo. 
         [0084]    On the contrary,  FIG. 7  shows an embodiment of an isolated ZETA converter. 
         [0085]    Those skilled in the art will appreciate that also a ZETA converter has, on the primary side of transformer T, the same electrical connection as a flyback converter. On the contrary, on the secondary side, the ZETA converter comprises an inductor Lz, a diode D 1  and two capacitors Co and Cz. Specifically, the first terminal of the secondary winding T 2  of transformer T is connected directly to the output, i.e. to line  106 . On the other hand, the second terminal of the secondary winding T 2  of transformer T is connected through capacitor Cz to the anode of diode D 1 . The cathode of diode D 1  is also directly connected to line  106 . Finally, the second output terminal, which represents ground GND 2 , is connected through inductor Lz also to the anode of diode D 1 . 
         [0086]    In this case as well, circuits  30  and  32  remain the same, and only the connection of circuit  32  on the secondary side T 2  of transformer T changes. 
         [0087]    Specifically, in the considered embodiment, the cathode of diode D 2  is still connected (directly) to the positive output terminal, i.e. line  106 , the anode of diode D 3  is connected (directly) to the second output terminal, i.e. ground GND 2 , and capacitor C 2  is connected (directly) to the anode of diode D 1 , i.e. between capacitor Cz and inductor Lz. 
         [0088]    Generally speaking, both circuit  30  and circuit  32  may comprise other components as well. 
         [0089]    For example,  FIG. 8  shows an embodiment of a flyback converter, the circuit  30  whereof comprises an RCD network, comprising capacitor C 1 , a diode D and a resistor R. 
         [0090]    Specifically, in the considered embodiment, diode D and capacitor C 1  are connected in series and resistor R and diode D are connected in parallel, so that the charging of capacitor C 1  is made easier by diode D, while the discharging of capacitor C 1  is slowed down by resistor R. 
         [0091]    Those skilled in the art will appreciate that such an RCD circuit may also be used in the other types of converters. 
         [0092]    Of course, without prejudice to the underlying principle of the invention, the details and the embodiments may vary, even appreciably, with respect to what has been described by way of example only, without departing from the scope of the invention as defined by the claims that follow.