Abstract:
Apparatus and methods for adjustment of spectral signal characteristics in polar modulators are described. A composite signal detection circuit is configured to detect when a signal trajectory of a composite signal provided to the polar modulator passes near the origin of a complex plane associated with the composite signal, and then adjusts the composite signal to pass through the origin. A spectral adjustment circuit is described to adjust AM and FM or PM components of the composite signal to reduce the deviation of an FM component of the composite signal.

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
     This application claims priority under 35 U.S.C. §119(e) to U.S. Provisional Patent Application Ser. No. 60/979,019, entitled POLAR MODULATION WITH IQ ZEROING, filed on Oct. 10, 2007. This application is also related to U.S. Pat. No. 6,985,703, entitled DIRECT SYNTHESIS TRANSMITTER, issued on Jan. 10, 2006, to U.S. Pat. No. 6,774,740, entitled SYSTEM FOR HIGHLY LINEAR PHASE MODULATION, issued on Aug. 10, 2004, to U.S. Provisional Patent Application Ser. No. 60/979,740, entitled FM PULSE SHAPING, filed on Oct. 12, 2007 and to U.S. patent application Ser. No. 11/369,897, entitled LINEAR WIDEBAND PHASE MODULATION SYSTEM, filed on Mar. 6, 2006. The content of each of these applications is hereby incorporated by reference herein in its entirety for all purposes. 
    
    
     FIELD OF THE INVENTION 
     The present invention relates generally to radio transmitters using polar modulation. More particularly but not exclusively, the present invention relates to apparatus and methods for reducing peak FM deviation without adversely affecting the spectrum of a composite transmit signal. 
     BACKGROUND OF THE INVENTION 
     Radio transmitters are used to generate the modulated signals required for wireless communications using modulation techniques such as QPSK, 8-P SK, 16-QAM, 64-QAM, and OFDM to vary the amplitude, phase, and/or frequency of the transmitter&#39;s RF carrier. 
     The modulated signal represents and conveys the message data consisting of in phase (I) and quadrature (Q) data streams. In practice, these data streams pass through digital filters that shape the resulting pulses and ultimately define the spectrum of the modulated transmit signal. A polar transmitter translates these I and Q data streams to equivalent amplitude (AM) and phase (PM) modulation signals. This allows these signals to be applied at more advantageous points in the transmitter, thereby increasing its efficiency. 
     The PM signal is applied to the RF carrier at a phase-locked loop (PLL). In practice, this is actually accomplished using the equivalent frequency modulation (FM) signal, which is easily found by differentiating the PM signal. Unfortunately, the differentiation process widens the bandwidth of the FM signal and also generates impulses. This is due to the fact that the phase jumps by as much as π whenever the transmit signal passes through or near the origin of the complex plane as shown in  FIG. 1 . The resulting FM impulses (that occur after differentiating the phase jumps), although infrequent, can be as strong as one-half of the data rate. 
     The FM signal&#39;s impulses and wide bandwidth present daunting challenges to the design of the polar transmitter. Any distortion of the FM signal alters the spectrum of the VCO output, elevates the noise floor around the transmit signal, and rotates the complex signal pattern. Practical circuits invariably reduce the bandwidth of the FM signal and degrade performance. More importantly, the VCO and PLL limit the peak FM deviation and corrupt the transmit output spectrum. 
     It would therefore be advantageous to reduce the peak FM deviation as well as the bandwidth of the FM signal. 
     SUMMARY 
     In one or more embodiments of the present invention, apparatus and methods for detecting and forcing the trajectory of a complex transmit signal to the origin when its amplitude drops below a threshold are described, providing potential advantages including reducing the mean value of the FM signal as well as reducing its wideband energy. 
     Additional aspects of the present invention are further described below with respect to the accompanying drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The following is a brief description of the drawings wherein: 
         FIG. 1  shows the complex signal trajectory for a WCDMA transmit signal; 
         FIG. 2  shows a diagram of simple polar transmitter; 
         FIG. 3   a  shows a diagram of a fractional-N phase locked loop (PLL); 
         FIG. 3   b  shows the noise contribution of a delta sigma modulator of the PLL as shown in  FIG. 3   a;    
         FIG. 4   a  shows a diagram of a phase/frequency modulator; 
         FIG. 4   b  shows the response of each of the modulation paths; 
         FIG. 5   a  shows a diagram of a dual port VCO; 
         FIG. 5   b  shows the response at the modulation port of the dual port VCO of  FIG. 5   a;    
         FIG. 6  shows the FM signal for a typical WCDMA transmit signal; 
         FIG. 7   a  shows one embodiment of a system for performing FM folding with extended AM in accordance with aspects of the present invention; 
         FIG. 7   b  shows a resulting FM signal provided by the circuit of  FIG. 7   a;    
         FIG. 7   c  shows a resulting AM signal provided by the circuit of  FIG. 7   a;    
         FIG. 8  shows example trajectories of a complex signal before and after IQ Zeroing to pass the signal directly through the origin, as implemented by one embodiment of the present invention; 
         FIG. 9   a  shows one embodiment of a system for implementing IQ zeroing in accordance with aspects of the present invention; 
         FIG. 9   b  shows Nyquist pulses combined with the I and Q signals, as may be used in embodiments of the present invention; 
         FIG. 9   c  shows the power spectral density (PSD) of FM/PM signals in the frequency domain before and after processing by addition of Nyquist pulses. 
     
    
    
     DETAILED DESCRIPTION 
     A simple diagram of a polar transmitter is shown in  FIG. 2 . The polar modulator synthesizes a transmit signal using direct phase modulation at the synthesizer and amplitude modulation at the variable gain amplifier (VGA) or power amplifier (PA). 
       FIG. 3   a  shows a fractional-N phase-locked loop (PLL) used to synthesize the radio frequency (RF) carrier signal. The PLL forms a feedback system that consists of a voltage-controlled oscillator (VCO), N counter, phase/frequency detector (P/FD), charge pump (CP), and integration filter (LPF). 
     The PLL uses negative feedback to force the phase of the feedback signal to track the phase of the reference signal. As a result, the VCO oscillates at a frequency given by
 
 f   VCO   =f   REF ( N+n )
 
where n represents the fractional value and N equals the integer value.
 
     The fractional-N phase-locked loop resolves fine frequency steps by modulating the value of Δn so that its average value satisfies 
             n   =         Δ   ⁢           ⁢   f       f   REF       =       1   M     ⁢       ∑     i   =   1     M     ⁢           ⁢     Δ   ⁢           ⁢     n   i                   
The ΔΣ modulator forms a sequence of Δn values with these important properties: 1) it responds to the input n quickly, 2) it possesses a resolution that improves with the number of samples, and 3) it concentrates quantization noise at high frequencies, near one-half the clock frequency.
 
     The quantization noise can be attributed to the integer nature of the feedback counter. It possesses a quantization error of ±½ around N or 
             Δ   =     1   N           
Assuming a uniform distribution of this error leads to the noise spectral density function described by
 
                 e   rms   2     ⁡     (   f   )       =     1     12   ⁢     N   2     ⁢     f   REF               
The ΔΣ modulator found in this polar transmitter shapes the quantization noise according to the transfer function
 
ΔΣ( z )=1 −z   −1 ) L  
 
where L is the order of the modulator. It in turn feeds the feedback counter, which acts a digital accumulator and reduces its noise-shaping effects. That is, the feedback counter operates in such a way that the current output phase depends on its previous output phase. As a result, the transfer function of the feedback counter or prescalar becomes
 
               P   ⁡     (   z   )       =     2   ⁢   π   ⁢       z     -   1         1   -     z     -   1                   
Combining the above equations shows that the noise at the output of the feedback counter equals
 
 n   2 ( f )= e   rms   2 ( f )[ΔΣ( f )] 2   [P ( f )] 2  
 
which simplifies to
 
                 n   2     ⁡     (   f   )       =       1   3     ⁢           π   2         N   2     ⁢     f   REF         ⁡     [     2   ⁢     sin   ⁡     (       π   ⁢           ⁢   f       f   REF       )         ]         2   ⁢     (     L   -   1     )                 
Ultimately, this noise must be attenuated by the loop filter and PLL transfer function to avoid excessive ΔΣ noise at the output of the PLL as shown in  FIG. 3   b.  
 
     To support wideband direct phase/frequency modulation, the fractional-N phase-locked loop adds a direct path to the VCO as shown in  FIG. 4   a . This modifies the VCO output to
 
ν out ( t )= A  cos [ω t+K   VCO ∫ν ctrl ( t ) dt+K   FM ∫ν FM ( t ) dt] 
 
where K VCO  and K FM  represent the sensitivity of the control port and the direct frequency modulation port, respectively. The FM signal also feeds the ΔΣ modulator and the feedback counter. This results in two paths for the FM signal as illustrated in  FIG. 4   b  and described by the transfer functions
 
               Δ   ⁢           ⁢   f     =           K   PD     ⁢     Z   ⁡     (   s   )       ⁢     K   V         sN   +       K   PD     ⁢     Z   ⁡     (   s   )       ⁢     K   V           ⁢   FM                   Δ   ⁢           ⁢   f     =         sNK   FM       sN   +       K   PD     ⁢     Z   ⁡     (   s   )       ⁢     K   V           ⁢   αFM           
where K PD  is the charge pump&#39;s gain, Z(s) is the impedance presented by the loop filter, K V  is the VCO&#39;s sensitivity at the tuning port, N is the value of the feedback counter, K FM  is the VCO&#39;s gain at the modulation port, and α is a scaling parameter. Ideally, these two functions combine to realize a flat response. That is, the ΔΣM path&#39;s response transitions smoothly to the VCO path&#39;s response and holds their combination at unity (0 dB). By its nature, the frequency modulation developed through the ΔΣ modulator is exact while, in contrast, the modulation formed at the VCO is sensitive to its gain K FM  and the accuracy of scaling parameter α.
 
     A key component of the direct phase/frequency modulator is the VCO shown in  FIG. 5   a . It uses complimentary MOS devices to replenish the losses in the LC resonator. The LC resonator consists of a differential inductor, coarse-tuning capacitors (not shown), and two variable capacitance structures based on accumulation-mode MOSFET devices. The accumulation-mode devices normally display an abrupt response, but impressing the large VCO signal across two back-to-back devices tends to linearize the response as shown in  FIG. 5   b . This is particularly important for wideband frequency modulation. 
     By design, signals applied to the control and modulation ports change the phase/frequency of the VCO output. Unfortunately, the VCO cannot discriminate between intended signals and noise. It therefore becomes important to minimize the noise as well as the sensitivity of these ports. Adding coarse-tuning capacitors to subdivide the VCO range lowers the sensitivity of the control port. Unfortunately, the nonlinear operations that form the FM signal can produce impulses as strong as one-half the FM data rate, as shown in  FIG. 6 . This is because the FM signal equals 
             FM   =         θ   ⁡     (   n   )       -     θ   ⁡     (     n   -   1     )           2   ⁢   π   ⁢           ⁢     T   R               
where T R  is the period of the phase (θ) and FM data. As such, a phase shift of ±π equals an FM deviation of ±1/(2T R ). In this example, the FM data rate equals 78 MHz, producing impulses approaching ±39 MHz.
 
     It&#39;s possible to cut these FM impulses in half by extending the AM signal since these impulses correspond to phase shifts approaching ±π. In practice, a phase shift of exactly ±π can be achieved by simply inverting the AM signal. This allows the strong FM impulses to map to a modified phase trajectory θ′ given by
 
θ′=θ± nπ 
 
where θ is the original phase and nπ represents the phase shift assigned to the AM signal. Alternatively, the strong FM impulses map to a modified FM′ given by
 
               FM   ′     =     FM   ±     1     2   ⁢     T   R                 
In essence, this operation folds strong FM impulses and effectively reduces the resulting or residual FM deviation.  FIG. 7   a  shows one embodiment of a folding operation in accordance with the present invention at the threshold TH 1 . This implementation inverts the polarity of the AM signal and removes the equivalent FM step from the FM signal whenever it exceeds the threshold. With the threshold set exactly to
 
               TH   1     =     1     4   ⁢     T   R               
the FM peaks fold over as shown in  FIG. 7   b . This effectively halves the FM range while it doubles the AM range (by making it bipolar).
 
     Conveniently, these strong FM impulses only occur when the AM signal moves towards zero. As a result, extending the AM signal actually smoothes the signal by removing inflections near zero as shown in  FIG. 7   c . In this example, the FM signal has been folded at about 76.4 μs, 77.1 μs, and 79.2 μs—flipping the polarity of the AM signal each time. 
     The strong FM impulses invariably occur when the composite transmit signal moves towards the origin (as mapped in the I/Q plane). This oftentimes corresponds to when the trajectory of the complex signal transitions between symbol points. Fortunately, the location of the symbol points is much more critical than the actual trajectory of the transmit signal. 
     In practice, it&#39;s possible to modify the signal&#39;s trajectory to provide some benefit but still route the transmit signal through the designated symbol points. This concept forces the signal trajectory that passes near the origin to actually pass through the origin as illustrated in  FIG. 8 . The result can be used to trigger FM folding, which flips the polarity of the AM signal and as a result better produces the FM impulse. 
     In one embodiment, the signal trajectory may be forced through the origin by using Nyquist pulses. These pulses may be added to the transmit signal whenever its complex trajectory (or AM signal) falls below a designated threshold TH 2 . In accordance with one embodiment, for ease of implementation the processing operates on I and Q data samples as shown in  FIG. 9   a . This implementation checks the I/Q values against the threshold TH 2 . If less than the threshold, separate and distinct pulses are added to the I/Q data. The pulse added to the I channel is scaled by the I data to force it to zero. Similarly, the pulse added to the Q channel is scaled by the Q data to drive it to zero. As a result, both I and Q signals intersect zero. 
     This processing alters the data streams as shown in  FIG. 9   b . As a result, the processing reduces the mean value of the FM signal (fewer FM peaks with generally less amplitude) and shapes the power spectral density of the AM and FM signals as shown in  FIG. 9   c.    
     In accordance with one embodiment, by design, the algorithm forces the FM impulse to occur when the AM signal equals zero. This may be accomplished by setting the PM sample—at the point n when the AM sample equals zero—to the same value as the previous PM sample with
 
 PM ( n )= PM ( n− 1)
 
where PM is equivalent to θ. Then, the derivative operation used to find the FM samples and defined by
 
 FM ( n )= PM ( n+ 1)− PM ( n )
 
properly positions the FM impulse at point n. In practice, setting PM(n)=PM(n−1) forces FM(n−1) to zero and usually pushes FM(n) to a value larger than threshold TH 1 . This consequently folds FM(n) and better realizes the FM impulse.
 
     In accordance with one or more embodiments, systems and methods for IQ zeroing may be used to advantageously decrease the wideband energy of the FM signal, which eases the design of the VCO and associated phase/frequency modulation system. 
     Some embodiments of the present invention may include computer software and/or computer hardware/software combinations configured to implement one or more processes or functions associated with the present invention, including those described above. These embodiments may be in the form of modules implementing functionality in software, hardware, and/or hardware software combinations. Embodiments may also take the form of a computer storage product with a computer-readable medium having computer code thereon for performing various computer-implemented operations, such as operations related to functionality as describe herein. The media and computer code may be those specially designed and constructed for the purposes of the present invention, or they may be of the kind well known and available to those having skill in the computer software arts, or they may be a combination of both. 
     Examples of computer-readable media within the spirit and scope of the present invention include, but are not limited to: magnetic media such as hard disks; optical media such as CD-ROMs, DVDs and holographic devices; magneto-optical media; and hardware devices that are specially configured to store and execute program code, such as programmable microcontrollers, application-specific integrated circuits (“ASICs”), programmable logic devices (“PLDs”) and ROM and RAM devices. Examples of computer code may include machine code, such as produced by a compiler, and files containing higher-level code that are executed by a computer using an interpreter. Computer code may be comprised of one or more modules executing a particular process or processes to provide useful results, and the modules may communicate with one another via means known in the art. For example, some embodiments of the invention may be implemented using assembly language, Java, C, C#, C++, or other programming languages and software development tools as are known in the art. Other embodiments of the invention may be implemented in hardwired circuitry in place of, or in combination with, machine-executable software instructions. 
     The foregoing description, for purposes of explanation, used specific nomenclature to provide a thorough understanding of the invention. However, it will be apparent to one skilled in the art that specific details are not required in order to practice the invention. Thus, the foregoing descriptions of specific embodiments of the invention are presented for purposes of illustration and description. They are not intended to be exhaustive or to limit the invention to the precise forms disclosed; obviously, many modifications and variations are possible in view of the above teachings. The embodiments were chosen and described in order to best explain the principles of the invention and its practical applications, they thereby enable others skilled in the art to best utilize the invention and various embodiments with various modifications as are suited to the particular use contemplated. It is intended that the following claims and their equivalents define the scope of the invention.