Abstract:
A receive signal processor jointly detects two or more symbols in a signal-of-interest in the presence of one or more other MIMO signals. The signal-to-interference-plus-noise ratio for each signal-of-interest is determined by computing per-subcarrier signal-to-interference-plus-noise ratios for a plurality of subcarriers allocated to the signals-of-interest, and computing a total signal-to-interference-plus-noise ratio for the subcarriers based on the per-subcarrier signal-to-interference-plus-noise ratios of the subcarriers. A controller determines one or more transmission formats for uplink transmissions based on the signal-to-interference-plus-noise ratios. The process of computing per-subcarrier signal-to-interference-plus-noise ratio reflects the amount of MIMO interference already cancelled or still remaining in the signal arriving at the joint detector.

Description:
BACKGROUND 
     The present invention relates generally to signal quality estimation in a mobile communication network and, more particularly, to signal quality estimation for the uplink in LTE systems. 
     Long Term Evolution (LTE) systems use single-carrier frequency-division multiple-access (SC-FDMA) for uplink transmissions. The use of single carrier modulation for the uplink is motivated by the lower peak-to-average ratio of the transmitted signal compared to conventional OFDM, which results in higher average transmit power and increased power amplifier efficiency. The use of SC-FDMA in the uplink, however, gives rise to an inter-symbol interference (ISI) in dispersive channels. It is important to mitigate the effects of ISI so that SC-FDMA can improve power amplifier efficiency without sacrificing performance. 
     Linear minimum mean square error (LMMSE) receivers in the base station (also known as an eNodeB) can suppress ISI using linear frequency domain equalization. LMMSE receivers are designed to maximize the signal-to-interference-plus-noise ratio (SINR) for each subcarrier component. Though LMMSE improves performance significantly beyond a simple match filtering receiver, further improvements in performance could be obtained with advanced receivers using techniques such as Turbo Soft Interference Cancellation (TuboSIC), or near maximum-likelihood detectors, such as a reduced state sequence estimator (RSSE), Serial Localization with Indecision (SLI) and Assisted Maximum Likelihood Detector (AMLD). These advance receivers are expected to achieve performance very close to the performance of a maximum-likelihood detector. 
     One problem encountered with the deployment of advanced receivers is obtaining reliable channel quality indication (CQI) estimation and modulation and coding scheme (MCS) selection. CQI estimates are used, for example, for link adaptation and scheduling in the uplink of LTE. Currently, there is no solution for CQI estimation for RSSE, SLI, MLD, or other ML, or near ML, receiver in the uplink in LTE systems. 
     SUMMARY 
     The present invention relates to the estimation of signal-to-interference-plus-noise ratios in multi-user receivers using ML or near ML detectors for the uplink of an LTE system. The embodiments of the present invention enable more accurate CQI estimation and MCS selection for multi-user single input, multiple output (MU-SIMO), single user multiple input, multiple output (SU-MIMO), and multiple user multiple input, multiple output (MU-MIMO) uplinks in LTE. The present invention can be applied to scheduling multiple uplink transmissions on the same frequencies (i.e., multi-user MIMO). 
     On exemplary embodiment of the invention comprises a method of determining a transmission format for uplink transmissions over a MIMO channel. One exemplary method comprises receiving two or more signals of interest over a multiple-input, multiple-output channel; iteratively detecting each signal-of-interest by successive interference cancellation and jointly detecting symbols within each signal-of-interest; and computing a signal-to-interference-plus-noise ratio for each signal-of-interest reflecting the remaining interference after cancellation of interference attributable to previously detected signals-of-interest. The signal-to-interference-plus-noise ratio may be computed by computing per-subcarrier signal-to-interference-plus-noise ratios for a plurality of subcarriers allocated to the signal-of-interest; and computing a total signal-to-interference-plus-noise ratio for the signal-of-interest based on the per-subcarrier signal-to-interference-plus-noise ratios of the subcarriers. The computed signal-to-interference-plus-noise ratios for the signals-of-interest are used to determine transmission formats for uplink transmissions. 
     Another exemplary embodiment of the invention comprises a receive signal processor for a communications device having one or more receive antennas. The receive signal processor comprises a detector to iteratively detect two or more signals-of-interest received over a multiple-input, multiple-output channel, and a signal quality estimator to compute a signal-to-interference-plus-noise ratio for the signal-of-interest reflecting the remaining interference after cancellation of interference attributable to the previously detected signals-of-interest. The detector jointly detects the symbols in each signal-of interest and then cancels the interference attributable to the detected signal-of-interest from the received signal until the last signal-of-interest is detected. The signal quality estimator is configured to compute per-subcarrier signal-to-interference-plus-noise ratios for a plurality of subcarriers allocated to the signal-of-interest; and compute a total signal-to-interference-plus-noise ratio for the subcarriers based on the per-subcarrier signal-to-interference-plus-noise ratios of the subcarriers. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  illustrates an exemplary block diagram of a MIMO transmitter. 
         FIG. 2  illustrates an exemplary receiver according to one embodiment of the present invention using a maximum likelihood (ML) or near ML detector to jointly detect a signal-of interest with one or more other signals. 
         FIG. 3  illustrates an exemplary method implemented by the receiver for computing a signal-to-interference-plus-noise ratio of the signal-of-interest detected using an ML or near-ML detector. 
         FIG. 4  illustrates a mapping function for translating a computed signal-to-interference-plus-noise ratio into a desired information rate and modulation scheme. 
     
    
    
     DETAILED DESCRIPTION 
     Referring now to the drawings,  FIG. 1  illustrates an exemplary MIMO transmitter  10  for generating SC-FDMA signals. An information bit stream is divided into two or more streams corresponding to the number of transmit antennas. Each bit stream is encoded in a channel encoder  12  (e.g., Turbo encoder) to produce a coded bit stream. The coded bit streams are modulated by modulators  14  (e.g., QAM) to generated time domain modulation symbol streams s n (0), s n (1), . . . , s n (K−1). Each modulated symbol stream is applied to a SC-FDMA transmitter  15  comprising a serial-to-parallel (S/P) converter  16 , a discrete Fourier transform (DFT) unit  18 , an inverse fast Fourier transform (IFFT) unit  20 , a cyclic prefix adder  22 , and a parallel-to-serial (P/S) converter  24 . Serial-to-parallel converter  16  converts a serial stream of time-domain modulated symbols s n (0), s n (1), . . . , s n (K−1) to a parallel substreams. DFT  20  converts the time-domain modulated symbols to frequency-domain symbols S n (0), S n (1), . . . , S n (K−1). As a result, each frequency-domain symbol is a function of all time-domain symbols in the input symbol stream. IFFT  20  applies an inverse Fourier transform to the frequency-domain symbols, cyclic prefix adder  22  adds a cyclic prefix to the IFFT output, and parallel-to-serial converter  24  converts the parallel symbols into a serial SC-FDMA signal stream. In frequency-selective channels, the time-domain symbols cannot be separated, interference-free, through linear equalization and IFFT. In this situation, ML or near-ML detectors that jointly detect the time-domain symbols s n (0), s n (1), . . . , s n (K−1) offer performance improvement. Although  FIG. 1  illustrates a separate encoder  12  and modulator  16  for each information bit stream, it will be appreciated that the information bit stream could be encoded and modulated prior to being divided. 
       FIG. 2  illustrates an exemplary receiver  100  according to one embodiment of the present invention. The receiver  100  includes front end circuits  110  and a receive signal processor  120 . The front end circuits  110  downconvert the received signal to baseband frequency, amplify and filter the received signal, and convert the received signal to digital form for input to the receive signal processor  120 . The main purpose of the receive signal processor  120  is to demodulate and decode a plurality of signals-of-interest. 
     In one embodiment of the invention, the receive signal processor  120  uses a multi-user detection (MUD) to detect a plurality of signals-of-interest. As used herein, the term multi-user includes the case where a single user is transmitting multiple streams, which from the perspective of the receiver, is the same as multiple users. Thus, the signals-of-interest may comprise signals from two or more different users, or multiple signals for a single user, or some combination thereof. 
     The exemplary receive signal processor  120  employs successive interference cancellation (SIC) to successively demodulate and decode the signals-of-interest contained in the received signal. The receive signal processor detects the signals-of-interest one at a time. After a signal is detected, the interfering signal can be recreated at the receiver using knowledge of the channel and subtracted from the received signal. This process is repeated successively, for each signal-of-interest, and progressively reduces the interference as each of the signals-of-interest is detected. Typically, the strongest signal is detected first and canceled from the received signal, which mitigates the interference for weaker signals. 
     The main functional components of the receive signal processor  120  comprise an interference canceller  122 , channel estimator  124 , impairment covariance estimator  126 , joint detector  128 , decoder  130 , signal generator  132 , and signal quality estimator  134 . The functional components of the receive signal processor  120  may be implemented by one or more microprocessors, hardware, firmware, or a combination thereof. 
     The received signal is input to the interference canceller  122 . The interference canceller  122  iteratively subtracts estimates of previously detected signals-of-interest from the received signal to generate a modified received signal for input to the joint detector  128 . During the initial iteration, the received signal is fed unchanged to the joint detector  128 . For each subsequent iteration, the detected signal from the previous iteration is fed back to the signal generator  132  to regenerate an estimate of the interference attributable to the detected signal. The interference estimate is subtracted from the received signal by the interference canceller  122 . This process is repeated until all of the signals-of interest have been detected. 
     The channel estimator  124  generates an estimate of the channel from one or more transmit antennas at a transmitting station (not shown) to one or more receive antennas (not shown) using any known channel estimation techniques. Typically, pilot symbols are used in the channel estimation process; however, data symbols could also be used as effective pilot symbols to improve channel estimation. The channel estimate produced from the pilot signal should be scaled appropriately to account for the power difference between pilot symbols and data symbols. The impairment covariance estimator  126  uses the channel estimates from the channel estimator  124  to estimate the covariance of the signal impairments, such as multi-user interference, self interference, other-cell interference, and noise, in the signals-of-interest. A new channel estimate and impairment covariance estimate is generated after each iteration. The channel estimates and impairment covariance estimates are input to the joint detector  128 , which uses the impairment covariance estimates along with the channel estimates to detect one of the signals-of-interest in the received signal. The channel estimates and impairment covariance are also used by the signal estimator  134  to generate estimates of the signals-of-interest. 
     The joint detector  128  preferably comprises an ML detector or near ML detector, such as a reduced state sequence estimator, SLI detector or AMLD detector. The joint detector  128  jointly processes symbols in each signal-of-interest contained in the received signal and generates received symbol estimates for each signal-of-interest. The received symbol estimates are demodulated to form received bit soft values that are then fed to a decoder  130 . The decoder  130  detects errors that may have occurred during transmission and outputs an estimate of a transmitted information sequence. 
     The signal quality estimator  134  estimates the signal-to-interference-plus-noise ratios (SINR) for the signals-of-interest. The SINR estimates may then be used to generate a channel quality indications (CQI), or modulation and coding scheme (MCS) selections. The CQI and/or MCS values may be reported to the transmitting station for link adaptation and/or scheduling. Therefore, reliable estimates of the SINR estimates are needed. Techniques are known for computing reliable SINR estimates for linear minimum means squared error (LMMSE) receivers for uplink in LTE systems. However, there are currently no known techniques for producing reliable SINR estimates for ML or near ML detectors for the uplink in LTE. 
     The signal quality estimator  134  according to embodiments of the present invention is able to produce reliable SINR estimates for single input/single output (SISO) and single input/multiple output (SIMO) systems.  FIG. 3  illustrates a method  150  according to one exemplary embodiment of the invention for generating SINR estimates for a signal-of-interest transmitted over an OFDM carrier. It is presumed that each signal-of-interest is transmitted from a single transmit antenna to one or more receive antennas. A single transmit antenna may be a single virtual transmit antenna which consists of a number of physical antennas. The signals-of interest can originate from multiple users or a single user. In the example, below, it is presumed that the signals-of-interest are received on multiple receive antenna. 
     The signals-of interest are detected successively by an SIC receiver as previously described (block  152 ) The SINR estimator  134  first computes a SINR estimate for each signal-of-interest. To compute SINR for a given signal-of interest, the SINR estimator  134  computes a per-subcarrier SINR estimate for each subcarrier allocated to the signal-of-interest (block  154 ). The SINR estimator then combines the per-subcarrier SINR estimates to obtain the final SINR estimate for the signal-of-interest (block  156 ). The SINR estimates for the signals-of interest can then be used to select a transmission format (MCS value) and/or to compute a CQI (block  158 ). 
     In one exemplary embodiment, the per-subcarrier SINR estimate for the nth signal-of-interest, denoted SIR[k] n , is computed according to:
 
SINR[ k]   n   =E   s,n   H   n   H   [k]R   w,n   −1   [k]H   n   [k]   Eq. (1)
 
where K is the total number of sub-carriers corresponding to the uplink spectrum allocation and k indexes the subcarriers, E s,n  is the symbol energy of the nth signal-of interest, H n [k] is the channel response vector collecting the frequency responses of the kth sub-carrier from the corresponding transmit antenna to all M receive antennas, H n   H  is the Hermetian transpose of the channel response vector, and R w,n [k] is the total impairment covariance at the kth sub-carrier. The channel response vector is given by:
 
                       H   n     ⁡     [   k   ]       =     [             H     0   ⁢           ⁢   n       ⁡     [   k   ]                   H     1   ⁢           ⁢   n       ⁡     [   k   ]               ⋮               H       M   -   1     ,   n       ⁡     [   k   ]             ]             Eq   .           ⁢     (   2   )                 
The impairment covariance matrix R w,n [k] may be computed according to:
 
                       R     w   ,   n       ⁡     [   k   ]       =         R   w     ⁡     [   k   ]       +       ∑     j   =     n   +   1         N   -   1       ⁢           ⁢       E     s   ,   j       ⁢       H   j     ⁡     [   k   ]       ⁢       H   j   H     ⁡     [   k   ]                     Eq   .           ⁢     (   3   )                 
where
 
               ∑     j   =     n   +   1         N   -   1       ⁢       E     s   ,   j       ⁢       H   j     ⁡     [   k   ]       ⁢       H   j   H     ⁡     [   k   ]               
is the covariance of the interference from other signals-of-interest that remain in the received signal, and R w,n [k] is the M×M matrix accounting for the noise and other cell interference covariance at the kth sub-carrier. In cases where the noise and other cell interference are uncorrelated across different antennas, R w,n [k] takes the form of a diagonal matrix.
 
     After the per-subcarrier SINR is obtained, the signal quality estimator  134  computes a total SINR for the signal-of interest based on the per-subcarrier SINR estimates. More specifically, the signal quality estimator  134  computes a per subcarrier capacity C k,n  for the subcarriers allocated to the signal-of-interest. The per-subcarrier capacity C[k] n  for a given subcarrier k is computed according to:
 
 C[k]   n =log(1+SINR[ k]   n )  Eq. (4)
 
The base of logarithm in the above capacity computation is 2 or other values. The per-subcarrier capacities C k  for the subcarriers allocated to the signal-of-interest are then summed and averaged by the signal quality estimator  134  to compute an average SINR given by:
 
                     C     AVG   ,   n       =       1   K     ⁢       ∑     k   =   0       K   -   1       ⁢       C   ⁡     [   k   ]       n                 Eq   .           ⁢     (   5   )                 
The average capacity C AVG  is then used to compute SINR MLD  of the signal-of-interest for a ML detector or near ML detector according to:
 
SINR MLD,n =exp( C   AVG,n )−1  Eq. (6)
 
     Combining Eqs. 1-6, the SINR MLD  for the signal-of-interest is given by: 
                     SINR     MLD   ,   n       =     exp   ⁢             (       1   K     ⁢       ∑     k   =   0       K   -   1       ⁢     log   ⁡     (     1   +       E     s   ,   n       ⁢       H   n   H     ⁡     [   k   ]       ⁢       (         R   w     ⁡     [   k   ]       +     
     ⁢                 ∑               ⁢     j   =     n   +   1                   ⁢     N   -   1         ⁢       E     s   ,   j       ⁢       H   j     ⁡     [   k   ]       ⁢       H   j   H     ⁡     [   k   ]             )       -   1       ⁢       H   n     ⁡     [   k   ]           )           )     -   1                 Eq   .           ⁢     (   7   )                 
The exp(x) and log(x) functions in Eq. (7) may, in some embodiments, be replaced by linear approximations or look-up tables.
 
     As previously described, the SINR MLD  may be used to generate a channel quality indication (CQI) and/or MCS (modulation and coding scheme) value to be reported to the transmitting station for link adaptation and/or scheduling. In SU-MIMO system, where the transmissions of the signals-of-interest originate from different users, the MCS/CQI to be reported for each signal-of interest can be obtained from a mapping function given by:
 
MCS n =MCSFormat(SINR MLD,n )  Eq. (8)
 
 FIG. 4  illustrates one exemplary mapping function for translating SINR MLD,n  to a target information rate and modulation scheme. If the modulation order is pre-determined, the MCSFormat function takes the information rate corresponding to the modulation order at the calculated SINR MLD,n  value. Alternatively, the MCSFormat function takes the highest information rate amongst the permitted modulation orders. That is, the modulation order is also dynamically and implicitly selected by the MCSFormat function. Such MCSFormat function can also be implemented as a look-up table stored in a retrievable storage/memory medium. More generally, to conserve MCS/CQI signaling bandwidth requirements, the MCSFormat transformation involves quantization of the SINR, which results in a look-up table with small number of entries.
 
     Further adjustments can be applied to SINR MLD,n  for link adaptation purposes. As non-limiting examples, such adjustments are typically applied to account for different quality of service requirements, losses induced by implementation imperfection, and channel quality variations within scheduling latency. To account for such variations, the mapping function for translating the SINR MLD,n  to MCS/CQI becomes:
 
MCS n =MCSFormat(SINR MLD,n −δ n )  Eq. (9)
 
where δ n  is the said SINR adjustment for the nth signal-of-interest. In one embodiment, the adjustments are dependent on the processing order in the SIC receiver to mitigate effects of potential error propagation. Decoding failures or errors in the signals that are processed earlier can increase the interference levels for the signals that are processed later. To minimize such error cases, it is advantageous to increase the error and interference resilience of the signals to be processed early in the SIC receiver by assigning decreasing values of the SINR adjustments (δ 0 ≧δ 1 ≧ . . . ≧δ N−1 ).
 
     For SU-MIMO systems, the N signals-of-interest are transmitted by a single user from an equal number of transmit antennas using K subcarriers. Depending on the system specifications, the N MCS/CQI values may be translated into N MCS/CQI values, one MCS/CQI value or L MCS/CQI values (with L&lt;N). 
     In the first case (separate MCS/CQI values), the N MCS/CQI values can be obtained by applying the MCSFormat mapping function to the N SINR MLD,n  values individually as shown in Eq. (9) or Eq. (10). 
     In the second case (only one MCS/CQI value), the SINR MLD,n  values may be combined into a single MCS/CQI value according to:
 
MCS=JointMCSFormat(SINR MLD,0 , . . . ,SINR MLD,N−1 )  Eq. (10)
 
If the modulation order is pre-determined, the JointMCSFormat mapping function takes the information rate corresponding to the modulation order at the calculated SINR MLD,n  value. Alternatively, full flexibility to use different modulation orders on different transmit antennas can also be supported. However, to conserve MCS/CQI signaling bandwidth, it may be a good engineering tradeoff to impose a single modulation order on the multiple transmit antennas corresponding to the same MCS/CQI signaling. In this case, the JointMCSFormat mapping function takes the highest information rate amongst the permitted modulation orders. That is, the modulation order is also dynamically and implicitly selected by the JointMCSFormat mapping function.
 
     One non-limiting example of the JointMCSFormat mapping function has the following form: 
                           M   ⁢           ⁢   C   ⁢           ⁢   S     =       ⁢     JointMCSFormat   ⁡     (       SINR     MLD   ,   0       ,   …   ⁢           ,     SINR     MLD   ,     N   -   1           )                       ⁢     SINR     MLD   ,     N   -   1         )               =       ⁢     Summary   (       MCSFormat   ⁢     (     SINR     MLD   ,   0       )       ,   …   ⁢           ,                         ⁢     MCSFormat   ⁢     (     SINR     MLD   ,     N   -   1         )       )     .                 Eq   .           ⁢     (   11   )                 
That is, the individual SINR MLD,n  values are first translated using the MCSFormat mapping function disclosed in the above. The translated values are then summarized into a single MCS/CQI value. If the modulation order is pre-determined, the individual MCSFormat mapping function takes the information rate corresponding to the modulation order at the calculated SINR MLD,n  value. Alternatively, full flexibility to use different modulation orders on different transmit antennas can also be supported.
 
     In the third case (L&lt;N MCS/CQI values), a single modulation order can be imposed on the multiple transmit antennas corresponding to the same MCS/CQI to conserve signaling bandwidth. The required translation procedure is as follows. A temporary modulation order is selected from the range of permitted modulation orders. The individual MCSFormat mapping functions are then evaluated based on the temporary modulation order selection. The final modulation order decision is set to the temporary modulation order that gives rise to the highest MCS/CQI level. 
     One non-limiting exemplary Summary function takes the form of weighting all translated values: 
                       M   ⁢           ⁢   C   ⁢           ⁢   S     =       JointMCSFormat   ⁢     (       SINR     MLD   ,   0       ,   …   ⁢           ,     SINR     MLD   ,     N   -   1           )       =       ∑     n   =   0       N   -   1       ⁢       w   n     ×     MCSFormat   ⁡     (     SINR     MLD   ,   n       )               ,           Eq   .           ⁢     (   12   )                 
where w n  is the weight for the nth antenna. One exemplary weight assignment is that w n =1/N for all n=0, . . . , N−1. Alternatively, uneven weighting may be applied per system setup requirements. To conserve signaling bandwidth, it may be necessary to quantize the summarized value:
 
     
       
         
           
             
               
                 
                   
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     In the third case, the system specifications require the N SINR MLD,n  values to be summarized into L MCS/CQI values (with L&lt;N). This is accomplished by breaking the N SINR MLD,n  values into L subsets. The SINR values in each subset are then combined using a JointMCSFormat translation function disclosed in the above. In combination with the teaching of minimizing error propagation, L different (and generally decreasing) SINR adjustments should be assigned for the L different SINR subsets: 
                       M   ⁢           ⁢   C   ⁢           ⁢     S   l       =     Quantization   (       ∑     k   =     n   l           n     l   +   1       -   1       ⁢           ⁢       w   n     ×     MCSFormat   ⁡     (       SINR     MLD   ,   k       -     δ   l       )           )       ,           Eq   .           ⁢     (   14   )                 
where {SINR MLD,n     l   , . . . , SINR MLD,n     l+1     −1 } is the list of SINR for the lth subset and δ 0 ≧δ 1 ≧ . . . ≧δ L−1 .
 
     In some embodiments, the MU-SIMO and SU-MIMO approaches can be combined to provide flexible balance of system and user throughput. Let M denote the total number of receive antennas at the base station and K denote the total number of sub-carriers allocated to the transmissions. Let U denote the total number of user terminals scheduled to transmit simultaneously on the allocated sub-carriers. For u=0, . . . , U−1, the uth UE transmits its signal with Nu antennas. As mentioned before, we only consider cases where the total number of uplink transmit antennas from all simultaneous scheduled UEs does not exceed to the total number of receive antennas. 
     The present invention provides a method and apparatus for easily computing the CQI for SLI, AMLD, or other ML, or near ML, receivers in the uplink of LTE. Thus, the present invention allows the base station to schedule a user to use a transmission rate that is more accurately reflecting the receiver capability, taking full advantage of advanced receiver performance. 
     The present invention may, of course, be carried out in other specific ways than those herein set forth without departing from the scope and essential characteristics of the invention. The present embodiments are, therefore, to be considered in all respects as illustrative and not restrictive, and all changes coming within the meaning and equivalency range of the appended claims are intended to be embraced therein.