Abstract:
In a CMOS bandgap reference circuit, the respective collectors of two lateral parasitic NPN transistors are connected to the two nodes of a current mirror. The emitter circuit of the first parasitic NPN transistor includes a resistor, whereby the base-emitter junction current densities of the parasitic NPN transistors are maintained at a preselected ratio. A second resistor common to the emitter circuit of both parasitic NPN transistors is provided, whereby ΔV BE  having a positive temperature coefficient and V BE  of the second parasitic NPN transistor having a negative temperature coefficient cancel one another. The temperature independent voltage across the common resistor and the base-emitter junction of the second transistor is buffered by a unity gain amplifier. The output of the unity gain amplifier is used to drive the parasitic NPN transistors and also is furnished as the reference voltage.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     This invention relates to CMOS circuits for generating a bandgap reference voltage, and more particularly to bandgap reference circuits that have reduced initial voltage reference error and temperature drift. 
     2. Description of Related Art 
     Reference voltage circuits are used by integrated circuit designers for many purposes, including analog to digital converters, regulated power supplies, comparator circuits, and some types of logic circuits. A particularly useful type of reference voltage circuit is the &#34;bandgap&#34; reference circuit, also known as the V BE  reference circuit, the theory of which is to generate a voltage with a positive temperature coefficient having the same magnitude as the negative temperature coefficient of V BE  ; then to add V BE  to the generated voltage to cancel the temperature dependency. 
     One type of parasitic npn bipolar transistors available from standard CMOS processes is a vertical transistor with its emitter, base and collector corresponding to, respectively, the source-drain n+ region, the p-well region, and the n- silicon substrate. The collectors of these parasitic vertical transistors are in the substrate, so that the transistors are suitable for use in a common collector configuration only. 
     One well known reference voltage circuit 10 which makes use of vertical parasitic transistors is illustrated in FIG. 1. VCC is applied at terminal 12, which corresponds to the substrate of the CMOS integrated circuit. Circuit ground is at terminal 14. Transistors 6 and 8 are parasitic NPN transistors, each of which uses the IC substrate for its collector, a P-well for its base, and an N-type drain/source region for its emitter. Resistors 20 and 22, which are the same value, are the load resistors for transistors 6 and 8 respectively. Resistor 24 is connected in the emitter circuit of transistor 6 to develop across it a temperature sensitive voltage. 
     The inputs of a differential amplifier 26 are connected across the equal valued resistors 20 and 22, and the output V REF , or reference voltage, is fed back to drive the bases of transistors 6 and 8. Due to this feedback, the potentials across the differential inputs at nodes 27 and 28 are equal (assuming amplifier 26 to be perfect, i.e. having infinite gain and input impedance). Even so, the current density in the emitter of transistor 6 is less than that of transistor 8 because of the voltage developed across resistor 24. Hence, transistors 6 and 8 exhibit different base-emitter potentials given by ##EQU1## wherein T is absolute temperature, k is the Boltzman constant, q is the charge of an electron, and I 8  /I 6 , A 6  /A 8  are the ratios of the current and emitter area of transistors 8 and 6 respectively. The quantity kT/q is also known as the &#34;volt-equivalent of temperature,&#34; and commonly represented by V T . 
     The difference in base-emitter potential ΔV BE  between transistors 6 and 8 appears across resistor 24 with a positive temperature coefficient. Since the current producing V R24  also flows through resistor 20, ΔV BE  having a positive temperature coefficient is imposed across resistor 22. Since resistors 20 and 22 are matched and the potential at nodes 27 and 28 maintained equal, a positive temperature coefficient attributable to ΔV BE  also is imposed across resistor 22. As V BE8  is of negative temperature coefficient, the one can be used to offset the other. 
     The value of ΔV BE  is set by establishing the respective emitter areas of transistors 6 and 8 at an appropriate ratio with the same I 6  and I 8 , in accordance with equation 1. Temperature compensation is achieved by adjusting value of R 20 , R 22  and R 24 . 
     Unfortunately, ideal CMOS amplifiers suitable for use as amplifier 26 are not available. Practical CMOS differential amplifiers have a temperature dependent input offset voltage that reduces the effectiveness of the bandgap reference circuit 10. The effect of the input offset voltage VOS on the bandgap reference circuit 10 is given by: ##EQU2## The input offset voltage of a CMOS differential amplifier typically is high; a value of greater than 2 mV is common. The ratio of (1+R20/R24) also is high; a value of 10 is common. Applying these common values, an error of 20 mV appears at the output of the amplifier 26, which does not permit the potential at nodes 27 and 28 to be maintained in equality. 
     Moreover, the input offset voltage is temperature dependent. The effect of this temperature dependency on the bandgap reference circuit 10 is given by the differential expression: ##EQU3## It will be appreciated that the offset voltage temperature dependency term δV OS  /δT is multiplied by the ratio (1+R 20  /R 24 ), which further degrades performance of the bandgap reference 10. 
     Several approaches have been taken in recognition of the performance limitations of the bandgap reference 10. One approach is to improve the performance of the differential amplifier used in the bandgap reference circuit 10, but this approach places significant restraints on the design of the amplifier 26. In any event, many of the causes responsible for the temperature dependent input offset voltage also are process sensitive. Another approach is typlified by U.S. Pat. No. 4,375,595, issued Mar. 1, 1983 to Ulmer et al. This and other such approaches increase circuit complexity and chip cost, however. 
     Recently, parasitic lateral NPN transistors have been used in the design of improved CMOS bandgap reference circuits. Two such circuits are disclosed in Degrauwe et al., &#34;CMOS voltage references using lateral bipolar transistors,&#34; in IEEE Journal of Solid State Circuits, Vol. SC-20, No. 67, Dec. 1985, pp. 1151-57. As shown in FIGS. 7(a) and 7(b) of the Degrauwe et al. article, these circuits use lateral bipolar transistors in combination with a current mirror, an output amplifier, and a voltage controlled current source. Unfortunately, the voltage controlled current source itself is fairly complex, being implemented by five additional resistors and an additional lateral transistor. Hence, the size of the bandgap circuit is increased. 
     SUMMARY OF THE INVENTION 
     An object of the present invention is to provide a relatively simple and cost effective CMOS bandgap reference circuit with improved temperature stability. 
     This and other objects are achieved by the present invention, a CMOS bandgap voltage reference circuit which uses two parasitic lateral bipolar transistors. The collectors of the lateral transistors are connected to one another A first resistor has one end connected to the emitter of one of the bipolar transistors. A second resistor has one end connected to the other end of the first resistor and to the emitter of the other bipolar transistor, and the other end connected to the ground potential. An amplifier, is connected to the collector of the other bipolar transistor, and its output is connected to the bases of the bipolar transistors. The potential between the amplifier output and ground potential is a reference potential. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     In the Figures, wherein like reference numerals indicate like parts: 
     FIG. 1 is a schematic diagram of a prior art bandgap reference circuit; 
     FIG. 2 is a generalized schematic diagram of a bandgap reference circuit in accordance with the present invention; 
     FIG. 3 is a detailed schematic diagram of the bandgap reference circuit of FIG. 2; and 
     FIG. 4 is a three dimensional view showing in cross section a portion of a parasitic NPN transistor used in the bandgap reference circuit of FIG. 2. 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENT 
     The reference voltage circuit 100 illustrated in FIG. 2 is suitable for fabrication with standard CMOS processes. Supply voltage VCC is applied at terminal 102, and circuit ground is provided at terminal 104. Transistors 106 and 108 are parasitic lateral NPN transistors, which include respective free collectors 126 and 128 and respective gates 122 and 124 which are biased as described below. A current mirror 110 comprising current sources 112 and 114 furnishes a current I112 to NPN transistor 106 and a current I114 to transistor 108, and maintains currents I112 and I114 equal. Resistors 116 is provided in the emitter circuit of transistor 106, and resistor 118 is provided in the emitter circuits of both transistors 106 and 108. A unity gain amplifier 120 has its input connected to the collector of transistor 108, and furnishes V REF  at its output 129. V REF  is fed back to the bases of transistors 106 and 108. 
     The operation of bandgap reference circuit 100 is as follows. Transistors 106 and 108 are driven by V REF . When transistor 106 pulls an incremental amount of current out of source 112 of current mirror 110, source 114 produces an equal increment of current into transistor 108. Hence, the current mirror 110 forces current I112 into the collector of transistor 106 and current I114 into the collector of transistor 108 to be equal. 
     Transistors 106 and 108 are fabricated with substantially identical diffusion profiles. Because of the difference in emitter area, the current densities across the base-emitter regions of transistors 106 and 108 are not equal. The different current densities result in different potentials across the base-emitter junctions of transistors and 108, given by ##EQU4## 
     The difference in base-emitter potential ΔV BE  between transistors 106 and 108 appears across resistor 116 for the following reason. Two branches connect the node at the bases of transistors 106 and 108 and the node 117, and the potential across the branches are the same. The potential across one of the branches is V BE108 . The potential across the other branch is the sum of the voltage drop across the resistor 116 (&#34;V R116  &#34;) and V BE106 . Node 117 forces V R116  +V BE106  to equal V BE108 , or 
     
         V.sub.R116 =V.sub.BE108 -V.sub.BE106                       (5) 
    
     Since applying equation 4 to transistors 106 and 108 yields the relationship ΔV BE  =V BE108  -V BE106 , it follows that V R116  equals ΔV BE . 
     The current producing V R116  also produces a voltage drop across resistor 118, which has a positive temperature coefficient as is evident from the sign of ΔV BE . The positive temperature coefficient attributable to ΔV BE  is imposed across resistor 118, and is effective for offsetting the negative temperature coefficient of V BE108 . 
     The value of V REF  is determined in accordance with the following expression: ##EQU5## where n is the ratio of emitter area of transistor 106 and 108. The appropriate ratio is established either by appropriately sizing the respective base-emitter regions or by connecting an appropriate number of identical transistors in parallel. 
     The temperature stability of bandgap reference 100 is given by: ##EQU6## Typically, δV BE118  /δT is about -2.0 mV/degree C and δV T  /δT is about +0.085 mV/degree C. The values of n and the ratio R118/R116 are selected to render δV REF  / δT zero, whereby a zero temperature coefficient is achieved. 
     The detailed schematic of the bandgap reference 100 shown in FIG. 3 is similar to the FIG. 2 schematic, except that the current mirror 110 and the amplifier 120 are shown in detail. Current mirror 110 is a CMOS current mirror of conventional cascode design. When parasitic NPN transistor 106 draws an incremental current through reference PMOS transistors 130 and 132, the source-drain voltage of transistor pairs 130,134 and 132, 136 are increased equally. Hence, transistors 134 and 136 produce an approximately equal increment of current into node 137. 
     To reduce offset in the current mirror 110, the mirror 110 is designed to be as symmetrical as possible, and the transistors 130, 132, 134 and 136 are designed as large area transistors. Transistors 130 and 134 are operated in the full saturation region to minimize the sensitivity to V cc  variation. 
     The amplifier 120 is a conventional two-stage source follower amplifier. The gate of the first stage PMOS transistor 138 is connected to the collector of transistor 108, and the drain is connected to ground. The base of the second stage, a conventional parasitic vertical NPN transistor 140, is connected to the source of transistor 138 and provides a low output impedance at its emitter, from which V REF  is taken. The collector of transistor 140 is in the substrate of the chip, which is connected to VCC. MOS transistor 139 is connected between VCC and the source of transistor 138 to provide a current path. The gate of transistor 139 is connected to the gate circuits of transistors 130 and 134 of the current mirror 110, which maintains the operation of transistor 139 in deep saturation. 
     For proper operation of the lateral transistors 106 and 108, VCC is applied to the substrate, which forms the collectors 126 and 128 of the associated vertical transistors, and the respective gates 122 and 124 are biased below their threshold voltage. The latter is achieved, for example, by connecting the gates 122 and 124 to ground 104, as shown, or to the emitters of transistors 106 and 108 respectively. 
     A transistor 200 suitable for use as transistors 106 and 108 is shown in FIG. 4. The transistor 200 is realized in a p-well CMOS process, although other CMOS processes are suitable as well. A p-well 204 is provided in n- substrate 202. A lateral parasitic NPN transistor is obtained from a concentric layout that includes a circular n+ diffusion region 206 which functions as an emitter, surrounded by a ring-like p- region 210 of the p- well 204 which functions as a base, surrounded in turn by a ring-like n+ diffusion region 212 which functions as a collector. Connection is made to the base 210 through a p+ diffusion region 208. A polysilicon gate 216 overlays base 210 and is insulated therefrom by gate oxide 218. A vertical parasitic NPN transistor is obtained from the emitter 206 and the substrate 202 using a region 214 of the p-well 204 between emitter 206 and substrate 202 as the base. Connection to region 214 is made through p+ region 208, and connection to the substrate 202 is made through n+ doped region 220. As the lateral transistor is more important than the vertical transistor when the parasitic transistor 200 is used as transistor 106 or 108, the length of base 210 (i.e. gate 216) is minimized and the perimeter-to-surface ratio of the emitter 206 is maximized. Contact is made to the various regions 206, 208, 212, 216 and 220 in any suitable manner, as is well known in the art. 
     Transistor 200 is operated as follows. Note that the collector 212 of the lateral transistor is not tied to the substrate, while the collector 220 of the vertical transistor is tied to the substrate. The lateral transistor is made operational by biasing the gate 216 far below its threshold voltage in order to create an accumulation layer in the region 210, thereby preventing MOS transistor operation between regions 206 and 212. Base 208, emitter 206, and collector 212 are suitably biased as discussed above. The associated vertical transistor is active since the substrate (i.e. collector 220) is tied to VCC. 
     Typical values for bandgap reference circuit 100 follow, for VCC equal to 5.0 volts and V REF  equal to 1.235 volts. Transistor 106 is laid out as eight individual transistors (n=8). Transistor 108 is laid out as an individual transistor. Transistor 108 and the individual transistors which combine to form transistor 106 are substantially identical. Transistor 140 is realized in such a way as to provide good drive capability. This is done by combining multiple individual transistors in parallel or by laying out the transistor with a large emitter area to boost the drive capability. Resistors 116 and 118 are p+ resistors of 1000 ohms and 7500 ohms respectively. Hence, the ratio R118/R116 is 7.5. Offset in the current mirror 110 is minimized by designing the mirror to be as symmetrical as possible. In addition, each transistor 130, 132, 134 and 136 is designed with a large area. The bandgap reference 100 requires no trimming. This is because there is no offset term in the reference generation circuit path. 
     While my invention has been described with respect to the embodiment set forth above, other embodiments and variations not described herein are to be considered within the scope of my invention. For example, my invention should not be limited by the specific type of transistor 200 used, or to any specific resistivity values and bias voltage values. These other embodiments and variations are to be considered within the scope of my invention, as defined by the following claims.