Abstract:
A digital audio broadcasting receiver comprises a phase error detector for detecting a phase error from data from a differential demodulator, an average value processing unit for determining the average value of phase errors, a memory for storing the phase errors of the carriers outputted from the phase error detector, and a phase error correcting unit which excludes a phase error whose sign is opposite to that of the average value among the phase errors stored in the memory, and determined the average value of phase errors again, thereby making it possible to obtain a phase error signal which is less affected by leakage from other carriers.

Description:
BACKGROUND OF THE INVENTION 
     The present invention relates to a digital audio broadcasting receiver in which each carrier is subjected to differential phase modulation and orthogonal frequency division multiplexing (OFDM). 
     As a system which permits transmission of digital data to a mobile object which is strongly affected by the problems of radio wave propagation, such as the multipath and fading, the orthogonal frequency division multiplexing (OFDM) transmission system is known, and the use of this system in broadcasting is under way. Its typical example is seen in digital audio broadcasting (DAB) which is set forth in ITU-R Recommendation BS.774. 
     FIG. 16 is a block diagram of a digital audio broadcasting receiver. 
     In the drawing, reference numeral  1  denotes an antenna;  2 , an RF amplifier;  3 , a frequency converter (MIX);  4 , a local oscillator (LO),  5 , an intermediate frequency amplifier (IF AMP);  6 , an orthogonal demodulator (DEMOD);  7 , an A/D converter;  8 , a synchronizing signal detector (synchronous detection);  9 , a synchronization control means;  10 , a complex discrete Fourier transform processing (hereafter referred to as “DFT”) means;  11 , a differential demodulator;  12 , a phase error detector;  13 , a frequency tuning control means;  14 , a Viterbi decoder;  15 , an MPEG audio decoder;  16 , a D/A converter;  17 , an audio amplifier; and  18 , a speaker. 
     In the receiver configured as described above, the broadcast wave received by the antenna  1  is amplified by the RF amplifier  2 , is subjected to frequency conversion by the frequency converter  3 , is subjected to removal of unwanted components such as adjacent channel waves and amplification by the intermediate frequency amplifier  5 , is subjected to detection by the orthogonal demodulator  6 , and is imparted to the A/D converter  7  as a baseband signal. 
     The signal sampled by the A/D converter  7  is subjected to DFT by the DFT means  10 , and the phase of each transmission carrier subjected to quadrature phase shift keying (QPSK) is detected. In the ensuing differential demodulator  11 , modulated phases of the same carrier of two transmitted symbols which are timewise adjacent to each other are compared, and processing (differential demodulation) for outputting a phase shift in the mean time is effected. The data subjected to differential demodulation is then outputted to the Viterbi decoder  14  in accordance with a rule on the order of carriers used in modulation on the transmitting side. 
     In the Viterbi decoder  14 , interleaving is canceled during the time spanning over the range of a plurality of symbols transmitted by the transmitting side, the data transmitted through convolutional coding is decoded, and correction of errors of data occurring on the transmission path is effected at that time. 
     In accordance with the provisions of the layer- 2  of ISO/MPEG 1 , the MPEG audio decoder  15  expands the compressed DAB broadcast audio data outputted from the Viterbi decoder  14 , and sends the same to the D/A converter  16 . The audio signal subjected to analog conversion by the D/A converter  16  is reproduced by the speaker  18  via the amplifier  17 . 
     Here, the synchronizing signal detector  8  detects the null symbol (the period during which no signal is present) by envelope detection, in the frame alignment signal included in the transmitted signal of DAB. This output serves as a timing signal by which DFT effected by the DFT means  10  through the synchronization control means  9  is executed correctly in synchronism with the transmission frame and each symbol of the signal. 
     The phase error detector  12  detects an error between an original phase point and the phase data of each carrier outputted from the differential demodulator  11 . That is, in DAB, if the frequency of the signal imparted to the orthogonal demodulator  6  is correct, the phase of the differentially demodulated data outputted from the differential demodulator  11  in correspondence with each carrier becomes substantially one of π/4, 3·π/4, 5·π/4, and 7·π/4. 
     Accordingly, if the data corresponding to each carrier is multiplied by 4 and the remainder is obtained with respect to 2π, this value becomes π if there is no error in the original data, and becomes a multiple of 4 of that value if there is a phase error in the original data, so that phase error detection is carried out. In practice, in the phase error detector  12 , the aforementioned operation is performed with respect to the data of the multiplicity of carriers, and the accuracy of detection is improved by averaging the results. 
     Since the phase error ε thus determined is an output from the differential demodulator  11 , the relationship of the following Formula (1) holds between an error ζ of the signal frequency at this time and the phase error ε: 
     
       
         ζ=ε/T   (1) 
       
     
     Here, T is a symbol period including a guard interval. 
     The frequency tuning control means  13  operates in such a manner as to cause the frequency error ζ of the baseband signal imparted from the orthogonal demodulator  6  to approach 0 by controlling the frequency of the intermediate frequency signal outputted from the frequency converter  3  by controlling the frequency of the local oscillator  4  in such a manner that this phase error ε becomes small. 
     As already described, the DAB signal is comprised of a multiplicity of carriers. To separate the carriers, DFT has an output characteristic shown in FIG. 17, and when the frequency is pulled in correctly, components from other carriers do not leak. 
     However, when the frequency is not pulled in correctly, components from other carriers leak, as shown in FIG.  18 . 
     Here, if there is no leakage from other carriers even if there is a frequency deviation, adjacent carriers s1and s2can be expressed by the following Formula (2): 
     
       
         s1=exp{j(2π(f0+Δf−n·fcc)t} 
       
     
     
       
         s2=exp{j(2π(f0+Δf−n·fcc) 
       
     
     
       
         (t+tsym)+θc+θn)}  (2) 
       
     
     where, f0: transmission frequency 
     Δf: frequency deviation 
     n: carrier number 
     fcc: interval between carrier frequencies 
     tsym: the period of one symbol 
     θn: (2N+1)p/4, N is an arbitrary integer 
     θc: 2π(f0−n·fcc)·tsym 
     Accordingly, the phase error from (2·.N+1)π/4 of the same carrier of adjacent symbols can be expressed by the following Formula (3): 
     
       
         θ=Δf·tsym  (3) 
       
     
     Hence, it can be seen that the phase error is proportional to the frequency deviation. 
     In practice, however, when the frequency has deviated, if there is leakage from other carriers, e.g., a frequency of −80 Hz, large variations appear in the differential modulated data, as shown in FIG.  19 . Here, the differentially demodulated data is divided into four quadrants of 0−π/2, π/2−π, π−3π/2, and 3π/2−2π, but there occurs data which enters adjacent quadrants as shown in FIG. 19, and the sign of the data which shifted to adjacent quadrants becomes opposite and such data constitutes a large phase error. Since erroneous data in which the sign of phase error is opposite is also used in averaging processing by the phase error detector, the detected phase error assumes a value smaller than a real value. 
     In addition, the greater the deviation of the frequency, the greater the leakage of components from other carriers, so that the variation becomes larger, and the data is located closer to the adjacent quadrants, with the result that the aforementioned error is liable to occur. For this reason, as for the frequency deviation and the average value of phase errors, the phase error becomes small starting from the frequency deviation of 70 Hz or thereabouts, where the frequency deviation and the average phase error cease to be proportional. For this reason, if the frequency deviation is large, there has been a problem in that it takes time in the pulling in of the frequency. 
     SUMMARY OF THE INVENTION 
     The present invention has been devised to overcome the above-described problem, and its object is to obtain a digital audio broadcasting receiver which is provided with a frequency control means for a local oscillator which is not affected by variations in the phase error due to the frequency deviation. 
     In the digital audio broadcasting receiver in accordance with the present invention, a DAB signal inputted from the antenna is subjected to OFDM demodulation by the DFT, the phase difference between two successive symbols on the same carrier is calculated by the differential demodulator, the deviation of the differentially demodulated data in an N-th quadrant from a (2N−1)π/4 radian is detected as the phase difference by the phase error detector, the phase errors of the carriers are averaged by the average value processing unit, the sign of the phase errors is detected by the sign determining unit, the phase error is corrected by excluding the effect of data which changed to an adjacent carrier by the phase error correcting unit in correspondence with the result of determination by the sign determining unit, and the frequency of the local oscillator is controlled by the corrected phase error. 
     In the digital audio broadcasting receiver in accordance with the present invention, a DAB signal inputted from the antenna is subjected to OFDM demodulation by the DFT, the phase difference between two successive symbols on the same carrier is calculated by the differential demodulator, the deviation of the differentially demodulated data in an N-th quadrant from a (2N−1)π/4 radian is detected as the phase difference by the phase error detector, the phase errors of the carriers are averaged by the average value processing unit, the sign of the phase errors is detected by the sign determining unit, and the frequency of the local oscillator is controlled by restoring the data which changed to an adjacent carrier by the phase error correcting unit in correspondence with the result of determination by the sign determining unit. 
     In addition, a DAB signal inputted from the antenna is subjected to OFDM demodulation by the DFT, the phase lo difference between two successive symbols on the same carrier is calculated by the differential demodulator, a phase rotation by a (2N−1)π/4 radian is imparted to the differentially demodulated data in an N-th quadrant by the phase rotating unit, the sign of imaginary parts of the data after the phase rotation is determined by the imaginary-part sign determining unit, addition is effected with respect to only the data whose signs of the imaginary parts are the same, the phase error detecting unit detects the phase error by excluding the effect of data which changed to an adjacent carrier, and the frequency of the local oscillator is controlled. 
     In addition, a DAB signal inputted from the antenna is subjected to OFDM demodulation by the DFT, the phase difference between two successive symbols on the same carrier is calculated by the differential demodulator, a phase rotation by a (2N−1)π/4 radian is imparted to the differentially demodulated data in an N-th quadrant by the phase rotating unit, the sign of imaginary parts of the data after the phase rotation is determined by the imaginary-part sign determining unit, addition is effected with respect to only the data whose signs of the imaginary parts are the same, the effect of data which changed to an adjacent carrier is restored by the phase error detecting unit, and the frequency of the local oscillator is controlled. 
     In addition, a DAB signal inputted from the antenna is subjected to OFDM demodulation by the DFT, the phase difference between two successive symbols on the same carrier is calculated by the differential demodulator, the deviation of the differentially demodulated data in an N-th quadrant from a (2N−1)π/4 radian is detected as the phase difference by the phase error detector, the phase errors of the carriers are averaged by the average value processing unit, the relative magnitude of leakage from another carrier is determined on the basis of output data from the differential demodulator, the average value of phase errors is corrected by the phase error correcting unit if the leakage from another carrier is large, and the frequency of the local oscillator is controlled. 
     In addition, a DAB signal inputted from the antenna is subjected to OFDM demodulation by the DFT, the phase difference between two successive symbols on the same carrier is calculated by the differential demodulator, the deviation of the differentially demodulated data in an N-th quadrant from a (2N−1)π/4 radian is detected as the phase difference by the phase error detector, the phase errors of the carriers are averaged by the average value processing unit, the variation of the differentially demodulated data is detected by the variation determining unit, the average value of phase errors is corrected if the variation is large, and the frequency of the local oscillator is controlled. 
     In addition, a DAB signal inputted from the antenna is subjected to OFDM demodulation by the DFT, the phase difference between two successive symbols on the same carrier is calculated by the differential demodulator, the deviation of the differentially demodulated data in an N-th quadrant from a (2N−1)π/4 radian is detected as the phase difference by the phase error detector, the phase errors of the carriers are averaged by the average value processing unit, an inclination of the magnitude of the phase error is detected by the inclination detecting unit, the average value of phase errors is corrected if the inclination is not in a converging direction, and the frequency of the local oscillator is controlled by the corrected phase error. 
     In the digital audio broadcasting receiver in accordance with the present invention, the phase error correcting unit handles the phase error whose sign is different from the sign of the average value of phase errors as being data in an adjacent quadrant since the phase difference in the differentially demodulated data has exceeded ±π/2, and determines that the data is erroneous. Hence, the phase error correcting unit corrects the phase error by effecting averaging with respect to only the phase errors whose sign agrees with the sign of the average value of phase errors. 
     In addition, in the digital audio broadcasting receiver in accordance with the present invention, in the restoration of the phase error by the phase error correcting unit, if the sign of the phase error is different from the that of the average value, the phase error is considered as being data in an adjacent quadrant since the phase difference in the differentially demodulated data has exceeded ±π/2. If the phase error is assumed to be q, the phase error correcting unit effects correction of θ−π/2 if the phase error is plus, and θ+π/2 if the phase error is minus. 
     In addition, in the digital audio broadcasting receiver in accordance with the present invention, if the absolute value is smaller between the absolute value of the sum of plus imaginary parts and the absolute value of the sum of minus imaginary parts, the phase error is ±π/2 or more, so that such data is handled as being data in an adjacent quadrant. Hence, since it is considered that the sign has been erroneous, the phase error correcting unit calculates imaginary parts/real parts of only the data whose absolute value is greater, and outputs the same as the phase error. 
     In addition, in the digital audio broadcasting receiver in accordance with the present invention, the phase error correcting unit calculates imaginary parts/real parts for plus imaginary parts and imaginary parts/real parts for minus imaginary parts, and the data which exhibits a greater absolute value between plus imaginary parts and minus imaginary parts is left as it is and is set as a phase error  1 . Meanwhile, in the case of data which exhibits a smaller absolute value is handled as data in an adjacent quadrant since the phase error has exceeded ±π/2, and it is considered that the sign of such data has been erroneous. Accordingly, if the phase error is assumed to be q, the phase error correcting unit effects correction of ±−π/2 if the imaginary parts are plus, and θ+π/2 if the imaginary parts are minus, thereby restoring the phase error to an original phase error as a phase error  2 . An average of the phase error  1  and the phase error  2  is used as the phase error. 
     In addition, in the digital audio broadcasting receiver in accordance with the present invention, if the leakage from another carrier becomes large, the phase error becomes smaller than a real value due to the leakage from another carrier, so that the phase error correcting unit provides processing for increasing the phase error, for example. 
     In addition, in the digital audio broadcasting receiver in accordance with the present invention, the greater the frequency deviation, the more the leakage from another carrier increases in the result of DFT, and the phase is also affected. The effect becomes large when the phase difference with respect to a neighboring carrier is ±π/2, but the phase difference with a neighboring carrier is not uniform. For this reason, the leakage from other carriers also changes. Therefore, variations occur in the result of differential modulation as shown in FIG.  19 . The greater the leakage from other carriers, the larger the variations, so that the variation of the data is calculated by the phase error correcting unit to determine the relative magnitude of leakage from other carriers. 
     In addition, in the digital audio broadcasting receiver in accordance with the present invention, the leakage component makes use of the fact that, between the region where the leakage from another carrier is large and the region where it is small, the signs of inclination of phase errors with respect to the frequency deviation are opposite. First, since feedback is provided to the local oscillator in such a manner that the phase difference approaches 0, the magnitude of the phase difference becomes small in the region where the leakage from other carriers is small. However, in the region where the frequency leakage is large, even if the phase error is small, the effect of leakage components from other carriers becomes small and the phase error approaches a real phase error, so that the phase error becomes apparently large. Accordingly, the leakage from other carriers is detected by performing a calculation in accordance with the following formula: 
     Δθ= (absolute value of the average value of current phase errors)−(absolute value of the average value of previous phase errors) If the sign of the previous errors and the sign of the current phase errors are the same, it is considered that the real phase error is approaching 0. As a result, if Δθ is plus, it can be determined that the phase error is becoming smaller than the real value due to the leakage from other carriers. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a block diagram illustrating the configuration of a digital audio broadcasting receiver in accordance with a first embodiment of the present invention; 
     FIG. 2 is a flowchart of processing by a phase error correcting section in accordance with the first embodiment; 
     FIG. 3 is a diagram illustrating the results of measurement of the phase error with respect to the frequency deviation in accordance with the first embodiment; 
     FIG. 4 is a block diagram illustrating the configuration of the digital audio broadcasting receiver in accordance with a second embodiment of the present invention; 
     FIG. 5 is a flowchart of processing by a phase error correcting section in accordance with the second embodiment; 
     FIG. 6 is a block diagram illustrating the configuration of the digital audio broadcasting receiver in accordance with a third embodiment of the present invention; 
     FIG. 7 is a flowchart of processing by a phase error correcting section in accordance with the third embodiment; 
     FIG. 8 is a block diagram illustrating the configuration of the digital audio broadcasting receiver in accordance with a fourth embodiment of the present invention; 
     FIG. 9 is a flowchart of processing by a phase error correcting section in accordance with the fourth embodiment; 
     FIG. 10 is a block diagram illustrating the configuration of the digital audio broadcasting receiver in accordance with a fifth embodiment of the present invention; 
     FIG. 11 is a flowchart of processing by a leakage component determining unit and a phase error correcting section in accordance with the fifth embodiment; 
     FIG. 12 is a block diagram illustrating the configuration of the digital audio broadcasting receiver in accordance with a sixth embodiment of the present invention; 
     FIG. 13 is a flowchart of processing by a leakage component determining unit and a phase error correcting section in accordance with the sixth embodiment; 
     FIG. 14 is a block diagram illustrating the configuration of the digital audio broadcasting receiver in accordance with a seventh embodiment of the present invention; 
     FIG. 15 is a flowchart of processing by an inclination determining unit and a phase error correcting section in accordance with the seventh embodiment; 
     FIG. 16 is a block diagram illustrating a conventional digital audio broadcasting receiver; 
     FIG. 17 is a conceptual diagram of DFT in a case where there is no frequency deviation; 
     FIG. 18 is a conceptual diagram of DFT in a case where there is a frequency deviation; 
     FIG. 19 is a diagram illustrating calculated values of differentially demodulated data owing to leakage from other carriers due to a frequency deviation; 
     FIG. 20 is a diagram illustrating calculated values of the phase error and the frequency deviation due to leakage from other carriers. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Hereafter, a specific description will be given of the embodiments of the present invention by referring to the drawings which illustrate its embodiments. 
     First Embodiment 
     FIG. 1 is a schematic block diagram illustrating a first embodiment of the present invention. In the drawing, reference numeral  1  denotes an antenna;  2 , an RF amplifier;  3 , a frequency converter (MIX);  4 , a local oscillator (LO),  5 , an intermediate frequency amplifier (IF AMP);  6 , an orthogonal demodulator (DEMOD);  7 , an A/D converter;  8 , a synchronizing signal detector (synchronous detection);  9 , a synchronization control means;  10 , a DFT means;  11 , a differential demodulator;  12   b , a phase error detector for detecting a phase error from (2N−1)π/4 of each carrier outputted from the differential demodulator  11 ;  13 , a frequency tuning control means;  14 , a Viterbi decoder;  15 , an MPEG audio decoder;  16 , a D/A converter;  17 , an audio amplifier;  18 , a speaker;  19 , a memory for storing an output of the phase error detector  12   b ;  21 , an average value processing unit for calculating the average value of outputs of the phase error detector  21 ;  22   a , a sign determining unit for determining the sign of an output from the average value processing unit  12   b ; and  23   a , a phase error correcting section for correcting the phase error by determining an error in the phase error from the result from the sign determining unit and by eliminating the error if the phase error is erroneous. 
     The data received by the antenna  1  is passed through the RF amplifier  2 , the frequency converter  3 , the intermediate frequency amplifier  5 , the orthogonal demodulator  6 , and the A/D converter  7 , and is subjected to complex discrete Fourier transform processing by the DFT means  10 . The signal demodulated by the DFT means  10  is subjected to differential demodulation by the differential demodulator  11 . Here, if the oscillation frequency of the local oscillator  4  has deviated, a phase rotation of a predetermined level or more occurs during the period of one symbol. 
     For this reason, the phase of the differentially demodulated data between the adjacent symbols deviates from (2N−1)π/4. The phase error detector  12   b  calculates the phase error θi of each carrier, and the memory  19  stores θi. 
     In addition, the averaging processing of phase errors detected by the phase error detector  12   b  is effected by the average value processing unit  21 . Since the data which shifted to adjacent quadrants is also included in this processing, the calculated value is smaller than a real phase error. However, it is considered from FIG. 20 that its sign does not change. 
     In the phase error correcting section  23   a , when the average value and the sign determined by the sign determining unit  22   a  are of the same data among the phase error data from the memory  19 , a control unit  24  turns on a switch  26 , and the average value of phase errors inputted from the memory  19  is calculated by an average value processing unit  27 . 
     FIG. 2 is a flowchart of processing by the phase error correcting section  23   a . Initialization is effected in Step  100 , and a determination is made in Step  101  as to whether or not the sign of the average value of phase errors and the sign of an i-th phase error are the same. If they are the same, in Step  102  the phase error θi are is added and the number n of the added carrier is incremented by 1. If they are not the same, i is incremented by 1 in Step  103 , and if i is greater than a value corresponding to the final data in Step  104 , a determination is made that it is the end of data, and the average value is calculated in Step  105 . 
     Through this processing, it is possible to detect a phase error without being affected by data which changed to adjacent quadrants. 
     FIG. 3 is a diagram illustrating the results of measurement of the phase difference due to the frequency deviation of differential decoded data in accordance with a conventional example and the first embodiment. In the first embodiment, it is possible to detect a phase error which is substantially proportional to the frequency deviation even up to a large frequency deviation, and the frequency tuning control means  13  changes the frequency of the local oscillator  4  by a frequency portion proportional to this phase error. 
     Second Embodiment 
     FIG. 4 is a schematic block diagram illustrating a second embodiment of the present invention, and the same reference numerals as those of FIG. 1 denote identical or corresponding portions, respectively. In the drawing, reference numeral  23   b  denotes a phase error correcting section which detects an error in the phase error in correspondence with the result of the sign determining unit and restores the phase error if in error. The phase error correcting section  23   b  is comprised of a control unit  24   b , a switch  26   b , the average value processing unit  27 , and a phase error restoring unit  28 . 
     Since the operation of the antenna  1  to the sign determining unit  22   a  is identical to that of the first embodiment, a description thereof will be omitted. Although the phase error data different in the sign from the output from the sign determining unit  22   a  is not used in the phase error correcting section  23   a  in the first embodiment, the second embodiment differs from the first embodiment in that the phase error data different in the sign is used. Namely, if the sign of the phase error data from the memory  19  is the same as the sign of the average value determined by the sign determining unit  22   a , the control unit  24   b  connects the switch  26   b  to the average value processing unit  27 , and connects the same to the phase error restoring unit  28  if the sign is different. The phase error restoring unit  28  effects the processing of θi ←θi−π/2 if the phase data θi≧0, and θi ←θi+π/2 if the phase data θi&lt;0, and outputs the result to the average value processing unit  27 . The average value processing unit  27  calculates the average value of the inputted phase errors. 
     FIG. 5 is a flowchart of processing by the phase error correcting section  23   b , in which eave denotes the average value of phase errors, θave&#39; denotes a correction value for the average value of phase errors, and θi denotes the phase error of an i-th carrier. 
     First, initialization is effected in Step  100   b , and if the sign of the average value θave of phase errors and the sign of an i-th phase error θi are the same, processing in Step  102  is effected, and if not, a determination is made in Step  106  as to whether or not the sign of the i-th phase error θi is plus. If plus, it is considered that the data of the quadrant which advanced by π/2 has changed, so that in Step  107  the phase error is corrected by using the phase which advanced by π/2 as a reference. If minus, it is considered that the data of the quadrant which lagged by π/2 has changed, so that in Step  108  the phase error is corrected by using the phase which lagged by π/2 as a reference. Subsequently, in Step  102 , the sum of averages of phase errors is calculated, and the carrier number i is incremented by 1. In Step  103 , if i is greater than the number of pieces of data, a determination is made that it is the end of data, and the average value is calculated in Step  104  and is outputted to the frequency tuning control means  13 . 
     Through this processing, it is possible to detect a phase error which is substantially proportional to the frequency deviation even up to a large frequency deviation, and the frequency tuning control means  13  changes the frequency of the local oscillator  4  by a frequency portion proportional to this phase error. 
     It should be noted that, in the phase error detection, the data in an N-th quadrant may be approximated by imaginary parts or real parts after imparting rotation by a −(2N−1)π/4 radian thereto. 
     Third Embodiment 
     FIG. 6 is a schematic block diagram illustrating a third embodiment of the present invention, and the same reference numerals as those of FIG. 1 denote identical or corresponding portions, respectively. In the drawing, reference numeral  23   c  denotes a phase error correcting section which is comprised of a comparator  34 , a switch  35 , and a divider  36 . 
     Reference numeral  29  denotes a phase rotating unit for detecting a phase error from (2N−1)π/4 in the data in the N-th quadrant of each carrier outputted from the differential demodulator  11 . Numeral  30  denotes an imaginary-part sign determining unit for determining the sign of imaginary parts of phase error in the output from the phase rotating unit  29 . Numeral  31  denotes a switch for changing over the output from the phase rotating unit  29  on the basis of the result of determination by the imaginary-part sign determining unit  30 . Numerals  32  and  33  denote adders connected to the switch  31 . 
     Since the operation of the antenna  1  to the speaker  18  is identical to that of the first embodiment, a description thereof will be omitted. 
     The signal demodulated by the DFT means  10  is subjected to differential demodulation by the differential demodulator  11 . Here, if the oscillation frequency of the local oscillator  4  has deviated, a phase rotation of a predetermined level or more occurs during the period of one symbol. For this reason, the phase of the differentially demodulated data deviates from (2N−1)π/4 . The phase rotating unit  29  rotates the output of the differentially demodulated data by a −(2N−1)π/4 radian. A phase deviation from a positive real axis after this operation becomes the phase error. 
     Hereafter, a description will be given of the operation with reference to the flowchart shown in FIG.  7 . First, initialization is effected in Step  200 , and a carrier is set in Step  201 . Next, in Step  202 , the imaginary-part sign determining unit  30  determines the sign of the imaginary parts of the output data from the phase rotating unit  29 , and changes over the switch  35 , and if the sign of the imaginary parts of the output data is plus, the output of the imaginary-part sign determining unit  30  is connected to the adder  32 , and calculations of Step  204  are performed. Meanwhile, if the sign of the imaginary parts of the output data is minus, the output of the imaginary-part sign determining unit  30  is connected to the adder  33 , and calculations of Step  203  are performed. 
     Next, if it is determined in Step  205  that it is the end of data, in Step  206  the comparator  34  of the phase error correcting section  23   c  compares the magnitude of the sum of imaginary parts between the outputs from the adder  32  and the adder  33 . If the absolute value of the sum of minus imaginary parts is greater than the absolute value of the sum of plus imaginary parts, the switch  35  is connected to the adder  33 , and the calculation of imaginary parts/real parts (Imm/Rem) is performed by the divider  36  in Step  207 . Meanwhile, if the sum of minus imaginary parts is not greater than the sum of plus imaginary parts, the switch  35  is connected to the adder  32 , and the calculation of imaginary parts/real parts (Imp/Rep) is performed by the divider  36  in Step  208 , and the result is outputted to the frequency tuning control means  13  as the phase error. As a result, it is possible to control the local oscillator  4  without being affected by the differentially demodulated data which changed to the data in adjacent quadrants due to leakage from other carriers. 
     Fourth Embodiment 
     FIG. 8 is a schematic block diagram illustrating a fourth embodiment of the present invention, and the same reference numerals as those of FIG. 6 denote identical or corresponding portions, respectively. In the drawing, reference numeral  23   d  denotes a phase error correcting section which is comprised of the comparator  34 , dividers  37  and  38 , a switcher  39 , an averaging unit  40 , and a restoring unit  41 . 
     Since the operation of the antenna  1  to the speaker  18  is identical to that of the first embodiment, a description thereof will be omitted. 
     The adder  32  adds complex data in a case where the sign of imaginary parts is plus, while the adder  33  adds complex data in a case where the sign of imaginary parts is minus. The divider  37  calculates imaginary parts/real parts for the adder  32 , while the divider  38  calculates imaginary parts/real parts for the adder  33 . The switcher  29  inputs to the averaging unit  40  the output from the divider whose absolute value of the sum of imaginary parts is greater, and inputs to the restoring unit  41  the output from the divider whose absolute value of the sum of imaginary parts is smaller. The outputs subjected to phase correction in the restoring unit  41  are then inputted to the averaging unit  40 . 
     FIG. 9 shows a flowchart. Since the processing from Step  200  to Step  205  shown in FIG. 9 is similar to that of the flowchart shown in FIG. 7 in accordance with the third embodiment, a description thereof will be omitted. In Step  206 , if imaginary parts are plus, the divider  37  calculates imaginary parts/real parts (Imp/Rep), and if the imaginary parts are minus, the divider  38  calculates imaginary parts/real parts (Imm/Rem). Next, in Step  207  the comparator  34  compares the absolute values of the sums of imaginary parts of the adders  32  and  33 . If the absolute value of the sum of minus imaginary parts of the adder  33  is greater than that of the adder  32 , θavep calculated on the basis of the imaginary parts of the adder  32  is erroneous, so that θavep is restored by processing in Step  209 . Meanwhile, if the absolute value of plus imaginary parts (imaginary parts of the adder  32 ) is greater than the absolute value of imaginary parts of the adder  33  in Step  207 , θavem is corrected by effecting processing in Step  213 . In Step  214 , the averaging unit  40  performs averaging processing with respect to the phase error which was not determined to be in error and the restored phase error. The result is outputted to the frequency tuning control means  13  as the phase error. 
     As a result, it is possible to control the local oscillator  4  without being affected by the differentially demodulated data which changed to the data in adjacent quadrants due to leakage from other carriers. 
     Fifth Embodiment 
     FIG. 10 is a schematic block diagram illustrating a fifth embodiment of the present invention, and the same reference numerals as those of FIG. 1 denote identical or corresponding portions, respectively. In the drawing, reference numeral  12   c  denotes a phase-error average value detector for detecting the average value of phase errors from (2N−1)π/4 of the carriers outputted from the differential demodulator  11 ;  42 , a leakage component determining unit which is connected to the differential demodulator  11  and estimates the magnitude of leakage from other carriers; and  23   e , a phase error correcting section which is connected to the leakage component determining unit  42  and corrects the average value of phase errors when the leak components are large. 
     Since the operation of the antenna  1  to the speaker  18  is identical to that of the first embodiment, a description thereof will be omitted. 
     The signal demodulated by the DFT means  10  is subjected to differential demodulation by the differential demodulator  11 . Here, if the oscillation frequency of the local oscillator  4  has deviated, a phase rotation of a predetermined level or more occurs during the period of one symbol. For this reason, the phase of the differentially demodulated data deviates from (2N−1)π/4. The phase-error average value detector  12   c  outputs the average value of phase errors of the carriers. 
     Hereafter, a description will be given with reference to the flowchart shown in FIG.  11 . 
     If the leakage component determining unit  42  determines in Step  300  that the leakage components from other carriers are large, the phase error correcting section  23   e  effects correction of phase error in Step  301 . 
     Hereafter, a description will be given of the processing in Step  301 . θp is a value of about 30 degrees or thereabouts and is a maximum value of phase error when there is an effect of carrier leakage. Here, the inclination of the phase errors when the effect of carrier leakage is large is about two times the inclination when it is small. By taking into consideration the fact that the phase error is proportional to this inclination and the frequency deviation, when the leakage from other carriers is large, correction is made so that the phase error approaches Formula (3) by using the following Formula (4): 
     
       
         θ′=3·θp/2−θ/2  (4) 
       
     
     where, θp: theoretically maximum value of the average value of phase errors 
     θ: output of the averaging processing means 
     θ: corrected phase error 
     By using this phase error, it is possible to control the local oscillator  4  without being affected by the differentially demodulated data which changed to the data in adjacent quadrants due to leakage from other carriers. 
     In addition, the operation in which the phase error increases may be added without complying with Formula (4). 
     For example, the coefficient of Formula (4) may be changed or replaced by a relatively large fixed value (e.g., 30 degrees which is a maximum value for a carrier in a case where there is leakage from other carriers). 
     Sixth Embodiment 
     FIG. 12 is a schematic block diagram illustrating a sixth embodiment of the present invention, and the same reference numerals as those of FIG. 10 denote identical or corresponding portions, respectively. In the drawing, reference numeral  43  denotes a variation detecting unit which is connected to the differential demodulator  11  and calculates the variation of the data;  23   e , a phase error correction unit which is connected to the leakage component determining unit and corrects the average value of phase errors when leakage components are large. 
     Since the operation of the antenna  1  to the speaker  18  is identical to that of the first embodiment, a description thereof will be omitted. 
     The signal demodulated by the DFT means  10  is subjected to differential demodulation by the differential demodulator  11 . Here, if the oscillation frequency of the local oscillator  4  has deviated, a phase rotation of a predetermined level or more occurs during the period of one symbol. For this reason, the phase of the differentially demodulated data deviates from (2N−1)π/4. The phase-error average value detector  12   c  outputs the average value of phase errors of the carriers. 
     In addition, if leakage from other carriers is large, the differentially demodulated data also varies. Accordingly, the variation s of the differentially demodulated signal is calculated in accordance with the following formula: 
     
       
         σ=Σ{(Rei−Reave) 2 +(Imi−Imave) 2 }  (5) 
       
     
     where, 
     Rei: real parts of i-th differential data 
     Reave: average value of real parts 
     Imi: imaginary parts of i-th differential data 
     Imave: average value of imaginary parts 
     Next, a description will be given of the operation of correction of the phase error with reference to the flowchart shown in FIG.  13 . In Step  302 , a determination is made as to whether or not the variation a is greater than a set value. If it is smaller, a determination is made that the leakage of components from other carriers is small, and the correction of phase error is not performed. 
     Meanwhile, if the variation a is greater than the set value, correction is made in Step  301 . The processing in Step  301  is similar to that in the fifth embodiment. 
     By using this phase error, it is possible to control the local oscillator  4  without being affected by the differentially demodulated data which changed to the data in adjacent quadrants due to leakage from other carriers. 
     In addition, the operation in which the phase error increases may be added without complying with Formula (4). 
     For example, the coefficient of Formula (4) may be changed or replaced by a relatively large fixed value (e.g., 30 degrees which is a maximum value for a carrier in a case where there is leakage from other carriers). 
     Seventh Embodiment 
     FIG. 14 is a schematic block diagram illustrating a seventh embodiment of the present invention, and the same reference numerals as those of FIG. 10 denote identical or corresponding portions, respectively. In the drawing, reference numeral  44  denotes an inclination determining unit which is connected to the phase-error average value detector  12   c  and monitors a change with time in the absolute value of the inclination of phase errors, and numeral  23   e  denotes a phase error correcting section which is connected to the phase-error average value detector  12   c  and the inclination determining unit  44 . 
     Since the operation of the antenna  1  to the speaker  18  is identical to that of the first embodiment, a description thereof will be omitted. 
     The signal demodulated by the DFT means  10  is subjected to differential demodulation by the differential demodulator  11 . Here, if the oscillation frequency of the local oscillator  4  has deviated, a phase rotation of a predetermined level or more occurs during the period of one symbol. For this reason, the phase of the differentially demodulated data deviates from (2N−1)π/4 . The phase-error average value detector  12   c  outputs the average value of phase errors of the carriers. 
     Next, a description will be given of the operation of the inclination determining unit  44  and the phase error correcting section  23   e.    
     In Step  303 , comparison is made between absolute values of the phase error detected this time and the phase error detected previously. Here, since feedback is provided to the local oscillator  4  in such a manner that the phase error approaches 0, the real phase error is smaller in the case of the phase error detected this time. 
     In practice, however, as can be appreciated from FIG. 20, when the frequency deviation becomes large and leakage components from other carriers become large, the detected phase error becomes smaller than the real phase error, whereas the closer to 0 the frequency deviation is, the larger the real phase error becomes than the detected phase error. Accordingly, if the current phase error is greater than the previous one in processing in Step  303 , it can be determined that the phase error is located in a region where the leakage components from other carriers are large. Hence, in Step  301 , correction of the phase error is effected in accordance with Formula (4) in the same way as in the fifth embodiment. 
     By using this phase error, it is possible to control the local oscillator  4  without being affected by the differentially demodulated data which changed to the data in adjacent quadrants due to leakage from other carriers. 
     In addition, the operation in which the phase error increases may be added without complying with Formula (4). 
     For example, the coefficient of Formula (4) may be changed or replaced by a relatively large fixed value (e.g., 30 degrees which is a maximum value for a carrier in a case where there is leakage from other carriers). 
     In accordance with the present invention, since the phase error calculated on the basis of the differential demodulation output data which changed to adjacent carriers due to the leakage from other carriers is excluded, the phase error due to the frequency deviation is prevented from becoming small, thereby making it possible to shorten the frequency pulling-in time. 
     In addition, since the phase error calculated on the basis of the differential demodulation output data which changed to adjacent carriers due to the leakage from other carriers is used after being corrected, the phase error due to the frequency deviation is prevented from becoming small, the average number of points is prevented from being reduced, and the variation of the phase error is prevented from becoming large, thereby making it possible to shorten the frequency pulling-in time. 
     Further, the phase error is determined after the differential demodulation output data which changed to adjacent carriers due to the leakage from other carriers is excluded. Hence, the phase error due to the frequency deviation is prevented from becoming small, thereby making it possible to shorten the frequency pulling-in time. 
     Further, the phase error is determined after the differential demodulation output data which changed to adjacent carriers due to the leakage from other carriers is excluded. Hence, the phase error due to the frequency deviation is prevented from becoming small, the number of averaging points is prevented from being reduced, and the variation of the phase error is prevented from becoming large, thereby making it possible to shorten the frequency pulling-in time. 
     Further, leakage components are determined from the differential demodulation output, and the average value of phase errors already calculated is corrected to a linear value which is substantially proportional to the frequency deviation, thereby making it possible to shorten the frequency pulling-in time. 
     In addition, the leakage from other carriers is detected in the region where the phase error becomes large with the lapse of time, and the average value of phase errors already calculated is corrected to a linear value which is substantially proportional to the frequency deviation, thereby making it possible to shorten the pulling-in time.