Abstract:
The design and performance of an analog cancellation system is presented. The system generates either narrow or wideband nulls in order to minimize the effect of interfering signals on a receiver. A microcontroller directs the detection and classification of the interfering signal relative to frequency, amplitude and modulation, such as pulse-width or continuous wave modulation. A sampled version of the interfering signal at frequency, fi, is phase-inverted, amplified, and vector-summed with the input signal stream to null the interfering signal at fi. The microcontroller also monitors and adjusts the cancellation systems&#39; circuit parameters to minimize any residual interfering signal at fi or respond to changes in the interference. The example system operates from 100-160 MHz, and can generate wideband nulls over a 5MHz bandwidth with a 15dB depth attenuation or narrowband nulls with a Q greater than 200, and with a null depth greater than 30dB.

Description:
STATEMENT OF GOVERNMENT INTEREST 
     The invention was made with Government support under contract No. F04701-93-C-0094 by the Department of the Air Force. The Government has certain rights in the invention. 
    
    
     FIELD OF THE INVENTION 
     The present invention is related to interference cancellation in communication systems. More particularly, the present invention is related to the implementation of an adaptive variable-bandwidth integrated interference cancellation system that minimizes the effects of undesired signals on receiver performance. 
     BACKGROUND OF THE INVENTION 
     A receiver can be subjected to undesired signals that are present over an operating bandwidth. Interfering signals can degrade the performance of wideband communication receivers. The undesired signals could be intentionally generated in order to jam or disrupt the receiver performance, or simply exist as a part of the surrounding signal environment. The interfering signals are classed as cosite or remote interferers. A cosite interferer is physically collocated with the receiver permitting a physical circuit connection from the interference generator to the receiver. A remote interferer is located far enough from the receiver to preclude a physical circuit connection. It is desirable to null interfering signals for improved performance. The design of the receiving antenna connected to the receiver and the physical separation between the interferer and the receiver antenna significantly affects the choice of the interference suppression system. 
     Often, adaptive antenna null-pattern generators are applied in order to minimize the effect of the unwanted signals on receiver performance. In other cases, the pattern of the antenna is not or cannot be adjusted. Instead, a sample of the interfering signal that is generated at a known location and having specific signal characteristics is obtained from an auxiliary antenna for the case of remote interference, or directly coupled from an interfering transmitter for the case of cosite interference. The system requires an auxiliary antenna, directional coupler, or multi-horn antenna to extract the interfering signal. The auxiliary antenna can be one of the horns in the main aperture of a multihorn array. However, some antenna systems have limited capabilities that adjust the antenna characteristics for obtaining a sample of the interfering signal through a known input location of the antenna system. One method of removing the interference is when the received signal is digitized and digital signal processing circuits can be used to filter out the undesired signals. Some receiver systems have limited processing capabilities for applying digital filtering techniques to a digitized version of the received signal. Instead all signals in the operating bandwidth are received, and adaptive filtering techniques are applied to these received signals to minimize the amplitude of any received but undesired signal. In this case, the undesired signal must first be detected, according to predefined criteria, and then isolated from the desired signals. 
     The general nulling function is well known and has been used in existing antenna systems. For Example, U.S. Pat. No. 5,729,829 discloses an interference mitigation method and apparatus for multiple collocated transceivers for band filtering of unwanted signals. Usually, a reference signal consisting of a non-coherent but correlated version of the undesired signal is obtained. The amplitude of the reference signal is equal to the amplitude of the interfering signal. The phase of the reference signal is set to 180° different from the interfering signal so that when the reference signal is reinjected back into the received signal, the undesired signal is cancelled in order to create a transmission null at the location of the undesired signal. When the receiver is collocated with the interference, a portion of the interference signal can be coupled from the transmission path by a directional coupler or another physical connection. This sampled signal is phase-shifted by 180° and vector-summed with the received signal. The 180° phase shift is produced by a vector modulation circuit. This vector sum is adaptively adjusted to produce a null at the frequency of the interfering signal. 
     One method uses transversal filters and mixers to generate the canceling signal. Another method uses a personal computer and a computation intensive routine to control a programmable transversal filter that detects the undesired signal. In these cases, the reference signals are obtained by coupling through additional antennas or by special connections to the interference source. Antenna arrays are used in communications systems. The signals from the array elements are vector summed together to produce the received signal. With adaptive control, the array can adjust the antenna pattern to minimize the effect of remote interference. The adaptive adjustment of the phase and amplitude weights of the array elements generates an antenna pattern null in the direction of the interfering signal. In other cases, a main antenna is combined with auxiliary antenna elements as a sidelobe canceller. In this case, the interfering signal is sampled by the broadbeam auxiliary antennas placed near the main antenna. The vector sum of the auxiliary antenna signals and the main antenna signal is adaptively processed to null the interference. The success of these adaptive antenna techniques depends on an ability to resolve the locations of the desired and interfering signals, and provide equalization to achieve effective interference over the required bandwidth. In many cases, sufficient space is unavailable to implement an array large enough to resolve the desired signals and remote interference. A wideband communication applications might preclude channelizing the operating bandwidth by a fixed channelization scheme or by a tunable bandpass filter, or by a lack of sufficient dynamic range to process large signal amplitudes. The above nulling systems use only relative signal power to determine whether a received signal is to be nulled. Adaptive filtering techniques could be applied to the unknown signals, but these techniques require initial conditions in the filter that depend on the characteristics of the received signals. 
     Usually prior cancellation methods require adjustment of the antenna pattern to create nulls for cancellation of unwanted signals, or external feeds containing unwanted signals that are then cancelled. In both cases, apriori knowledge is required. These prior methods typically use a narrowband tunable bandpass filter as a preselector at the front end of the receiver. The front-end preselector has a disadvantage in a wideband communications receiver. The narrowband preselector would filter out most of the desired signal along with an interfering signal. Series tunable band-notch filters could be placed before the receiver. The bandpass and bandnotch filtering methods are serial in-line processes that reduce the reliability of the receiver. When the tuning mechanism in the preselector fails, the filter may lock at one center frequency, other signals cannot be received. The disabled filter would then significantly and permanently degrade receiver performance in part of the passband. These and other disadvantages are solved or reduced using the invention. 
     SUMMARY OF THE INVENTION 
     An object of the invention is to provide cancellation of unwanted received signals received by a communication receiver 
     Another object of the invention is to provide cancellation of unwanted signals having predetermined frequency, amplitude and modulation criteria. 
     Yet another object of the invention is to provide scanning by searching selected frequencies for unwanted signals having predetermined frequency, amplitude and modulation criteria and to cancel the located unwanted signals to result in desired received signals. 
     Still another object of the invention is to provide an adaptive variable bandwidth cancellation system for isolating and canceling unwanted signals having predetermined frequency, amplitude and modulation criteria. 
     The present invention is directed to a microcontroller based adaptive variable-bandwidth cancellation system for use in a wideband communication receiver system. The cancellation system is placed in parallel with and becomes part of a receiver. The use of a microcontroller allows for flexibility in defining the characteristics of the interfering signal. The preferred cancellation system provides narrowband and wideband cancellation nulls for canceling unwanted interfering signals. The limitation on null depth is caused by the finite resolution of the phase-shift transmission lines and attenuation steps. The signals within the scanned frequency bandwidth are detected in a detection path and parameterized according to frequency, amplitude or modulation, such as pulse-width modulation or continuous wave modulation. These characteristics are then compared against the definition of an undesired signal that is stored in the microcontroller. When an undesirable signal is detected, a tunable reference path is set so as to cancel the undesirable signal from the received signal and so as to reduce the undesirable signal detected signal. Iterations of detection and cancellation achieve desired cancellation of the unwanted signal using adaptive cancellation. The detection path is tunable for scanning across step bandwidths for detecting unwanted signals of interest. Once an undesirable signal is located at a particular frequency location, the tunable reference path is tuned to that particular frequency location to isolate the undesired signal that is then inverted and added to the composite receive signal to cancel the unwanted signal from the composite receive signal to provide only desired received signal with the detected unwanted signal canceled. The reference path serves to isolate an undesired signal from the desired signals, and then serves to amplify and shift the phase of the undesired signal for nulling summation with the original received signal that is delayed for coherent nulling. Once an undesired signal has been detected, the microcontroller sets the values of the reference path components according to a predetermined look up table. The undesired received signal is continuously fed into the detection path for monitoring the effect of the cancellation and when further cancellation is needed, the reference path is appropriately tuned to remove the undesired signals. The microcontroller adaptively continuously scans the receiver bandwidth and monitors the detected signal from the detection path searching for unwanted interfering signals, and characterizes the detection signals, and then tunes both reference path and detection path circuit parameters to maximize detection of unwanted signals to minimize the amplitude of an undesired signal in the surviving received signal. The microcontroller-based system is preferred for signal detection and evaluation. Undesired signals are detected and isolated at a location internal to the cancellation circuitry. Tunable bandwidth bandpass filters in the detection path and reference path are used to generate wideband or narrowband nulls depending on the signal characteristics. The controller continuously monitors the cancellation result by sensing the detection signal from the detection path, and adaptively minimizes any residual of the interfering signal. 
     The use of a programmable microcontroller allows for flexibility in the detection and classification of interfering signals. In the preferred form, the microcontroller searches for either narrowband or wideband signals. Detection threshold amplitudes can also be varied as a function of frequency. The cancellation system can be used in space or airborne applications wherein weight, size, and power are prime considerations. The system also has applications in the commercial sector where receivers, such as GPS receivers, are used near emitters at the same frequency or at multiple harmonics of television or radio station frequencies. The microcontroller can be efficiently programmed without floating-point mathematics, matrix inversion, or other higher mathematical functions. The controller can be programmed so that different classes of signals are cancelled depending on signal parameters, such as frequency, pulse-width, and amplitude a stand alone configuration with external controls. In the event of system failure, the cancellation function can be disabled without affecting reception of the received signal by the operating receiver. 
     The microcontroller enables flexible programming and adaptive control allowing for compensation of component performance drift over lifetime and environments in which the system cannot be reached for manual repair or replacement of parts. Flexible control of the signal detection system allows for a relatively large detection range of over 60 dB using amplitude detection techniques. The null also can be located within a wide bandwidth according to a relatively coarse calibration table. The microcontroller can then measure the null efficiency and adjust the cancellation performance to improve the null. In an exemplar form, the system can generate 5 MHz wideband nulls with 15 dB in cancellation or narrowband bandwidth nulls with 30 dB in cancellation over a 100 MHz to 160 Mhz operating bandwidth. These and other advantages will become more apparent from the following detailed description of the preferred embodiment. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1A is a schematic of an analog nulling and detection circuit. 
     FIG. 1B is a block diagram of a digital control circuit. 
     FIG. 2 is a schematic of a tunable bandwidth bandpass filter. 
     FIG. 3 is an adaptive nulling flow diagram. 
     FIG. 4A is a graph of a narrowband amplitude bandpass filter transfer function. 
     FIG. 4B is a graph of a narrowband phase bandpass filter transfer function. 
     FIG. 5A is a graph of a wideband amplitude bandpass filter transfer function. 
     FIG. 5B is a graph of a wideband phase bandpass filter transfer function. 
     FIG. 6A is a graph depicting a wideband received signal. 
     FIG. 6B is a graph depicting the wideband received signal with signal cancellation. 
     FIG. 7A is a graph depicting a narrowband received signal. 
     FIG. 7B is a graph depicting the narrowband received signal with signal cancellation. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     An embodiment of the invention is described with reference to the figures using reference designations as shown in the figures. Referring to FIG. 1A, an analog nulling and detection circuit receives and processes an input signal  10 . The input signal  10  contains both desired and undesired signals that are collected by an input antenna. The input signal  10  is received and communicated through a delayed path consisting of a time delay  12  to provide a delayed input signal to a directional coupler  14 , and communicated through a reference path consisting of a first low noise amplifier (LNA)  18  providing a reference bandpass filter input signal  19 , a reference bandpass filter (BPF)  20  providing a reference BPF output signal  21 , a second reference LNA  22 , reference phase shifter  24  that may be a switched delay line, and a variable gain amplifier (VGA)  26  providing a reference signal communicated to the directional coupler  14 . The directional coupler  14  receives the delayed input signal from delay  12  and receives the reference signal from the VGA  26  and couples these two signals together to provide an output signal  16 . The output signal is communicated through a detection path consisting of a detection LNA  28 , a detection BPF  30 , a detection VGA  32 , a square law diode  34 , and a detection lowpass filter  36  that provides a baseband detection output signal  38 . The detection BPF  30  is used to sweep frequency bands of interest for detection of undesired signals within each stepped frequency band. The baseband detection output  38  indicates the presence of desired and undesired signals within the frequency band selected by detection BPF  30 . The output of the 10 dB directional coupler  14  is firstly communicated to the output  16 . A sensed portion of the output  16  is communicated through the LNA  28 , the tunable BPF  30 , the VGA  32 , square-law diode  34  and LPF  36  of the detection path in order to detect the presence and frequency of an undesired signal. The LNA  18 ,  22 , and  28  may be CLC 449  current-feedback op-amps to provide gain and high input impedance. In order to minimize loss in the delay path, the input signal  10  and the output signal  16  are connected to high input-impedance amplifiers  18  and  28 . Furthermore, if the active components in the reference and detection paths are disabled, the paths do not load the delay path. 
     The detection VGA  32  and detection BPF  30  are respectively controlled by a D 1  analog control signal and a D 2  digital control signal. The reference BPF  20  is controlled by D 3  and D 4  analog control signals. The reference phase shifter  24  and VGA  26  are respectively controlled by digital control signal D 5  and analog control signal D 6 . The D 1  and D 2  signals are used to control the operation of the detection path consisting of elements  28 ,  30 ,  32 ,  34  and  36 . The D 3 , D 4 , D 5  and D 6  signals are used to control the operation of the reference path consisting of elements  18 ,  20 ,  22 ,  24  and  26 . D 1  is an analog gain setting signal. D 2  is a digital bandpass tuning signal. D 3  is an analog BPF bandwidth tuning signal. D 4  is an analog BPF center frequency setting signal. D 5  is a digital phase shift adjustment signal. D 6  in an analog gain setting signal. In operation, the delay  12  is matched to the delay through the reference path, so that input signal  10  and the reference signal from the VGA  26  are time synchronized upon additive reception by the direction coupler  14  providing minimum insertion losses to the output signal  16  that is the sum of the input signal  10  and the reference signal from the VGA  26 . The reference path serves to shift by 180° the input signal  10  with appropriate gain and phase so as to cancel unwanted interference signal within the input signal  10  thereby converting the input signal  10  into the output signal  16  having canceled interference. For minimizing insertion losses, the directional coupler  14  maybe a conventional splitter that is reversibly connected so that the delayed input signal is received at the splitter output, the reference input from VGA  26  is received at the −10 db output of the splitter  14 , and the output signal  16  is provided at the input of the splitter. The delayed signal from the delay  12  and the reference signal from the reference path are set to have matched delays for coherent combining in the directional coupler  14 . The delay  12  may be, for example, 17 ns to match the delay through the reference path. The reference signal transits the reference path, while the delayed signal passes through the 17 ns delay line  12  and into the nominal output port of the 10 dB directional coupler  14 . The nominal output port and coupling ports are used as inputs, while the nominal input port is used as an output. In this manner, isolation is provided between the reference and delay signal paths, along with approximately 2.5-dB less insertion loss over an alternative conventional 3-dB coupler. 
     The output of the reference BPF  20  is then fed into the phase shifter  24 . The phase shifter, for example, may consist of two amplifier channels, not shown, one of which can be chosen at a time. One amplifier channel is designed as an inverting amplifier while the other channel is designed as a noninverting amplifier. The output of this circuit is then fed to a switched delay line, not shown. Any or all of the six delay lines can be switched into the reference path as necessary. The delay lines have nominal electrical lengths of 90°, 45°, 22°, 11°, 60° and 3° at 100 MHz. Thus, a phase shift of 360° can be achieved through the combined use of one of the channels and the switched delay line. The output of the phase shifter  24  is then fed into the VGA  26  that may be a CLC 522  VGA. The VGA is used to adjust the reference path signal level. During the detection signal scanning, the VGA  26  is set to maximum attenuation. Additional gain stages, not shown, may be used to increase the reference path signal level for injection into the coupled port of the 10 dB directional coupler  14 . All of the signal levels, even at the input to 10 dB directional coupler  14 , are considerably below the 1 dB compression and third order intermodulation intercept points of the active components. 
     Referring to FIGS. 1A and 1B, and more particularly to FIG. 1B, a digital control circuit is used to control the operation of the analog nulling and detection circuit of FIG. 1A, so that interference in the reference path is canceled from the input signal  10  as the output signal  16 . The digital control circuit receives the baseband detection output  38  and provides the control signals D 1 -D 6  to the detection and nulling circuits. The baseband detection output  38  is received and converted by a signal amplitude analog to digital converter  40  providing a sample and hold signal that includes a sign value and an amplitude value. The baseband detection output  38  is also received by a threshold comparator  42  providing a threshold comparator output to a pulse width detection circuit  44  and to a continuous wave (CW) detection circuit  46 . The comparator  42  provides an output that indicates if a signal has been detected. The detection circuits  44  and  46  provide respective CW and pulse outputs to a selector  48  that selects the CW or pulse output from the CW and pulse circuits  46  and  44 , respectively, and then communicates the selected output to a microcontroller  50 . The microcontroller  50  received the sampled output from the converter  40 , the selected output from the selector  48  as inputs and provides control signals to the comparator  42 , converter  40  and selector  48 , as well as providing the digital control signals D 1 -D 6 . The microcontroller  50  controls the operation of the comparator  42  by providing a threshold level signal that is varied depending on how small a detected signal is to be detected above a noise floor. The digital control circuit implements an automated analog cancellation or nulling method that generates narrow and wideband nulls in order to minimize the effect of undesired interference signals on a received signal  10 . The control circuit detects and isolates the undesired signal by controlling the operation of the detection path and reference path. The digital control circuit does not rely on apriori information about the undesired signal. The control circuit allows all desired input signals of input  10  to pass through to the output  16 , while nulling, that is canceling, the undesired signals using the reference path. 
     The baseband detection signal  38  is communicated to the cw detection circuit  46  and to the pulse detector circuit  44 . These detectors  44  and  46  provide an active output if the baseband detection signal exists above the threshold value of the comparator  42  as controlled by the microcontroller  50 . If the detected signal exceeds the threshold value, the detected signal is categorized as a desired or undesired signal, and as a pulsed or as a cw signal, and the amplitude and sign of the signal from the converter  40  as well as the current values of the detection control signal D 1  and D 2  for the detection signal are then used to determine through look up tables, the values of the nulling control signal D 3 , D 4 , D 5  and D 6  used in the reference path for cancellation. Additionally, microcontroller  50  can determine if a pulse signal is less or greater than a predetermined value, and any pulsed signal longer or shorter than this value, respectively, and any cw signals are designated as undesired signals can be nulled as well. 
     The null generation is achieved by summing outputs of the delay path and the reference path at a given frequency or over a given frequency band. The undesired signals in the delay path and the reference path must have the same amplitude within an amplitude offset but phase shifted by 180° within a phase offset. The reference path has a frequency response defined by components within the reference path including the BPF  20 . The BPF  20  in the reference path is required to isolate the undesired signal from the desired signal within a given bandwidth but having associated group delays. The BPF  20  has an amplitude response that is not flat over the operating bandwidth and the phase length in the reference path is larger than that of the delay path. The BPF  20  affects the amplitude and phase of signals in the band so that phase shifting by shifter  24  and variable gain by the VGA  26  is required to match the isolated signal in the reference path to the unwanted signal in the delay path for cancellation. A detailed circuit diagram of the tunable bandwidth BFP  20  is shown in FIG.  2 . 
     Referring to FIGS. 1A,  1 B and  2 , and more particularly to FIG. 2, the BPF  20  provides tunable filtering. The BPF input  19  is communicated through a first DC block capacitor  60  to a MV209 varactor diode  62  functioning as a tuning element. The diode  62  is connected to an inductor  64  and a second DC blocking capacitor  66 . The inductor  64  is connected to a bypass capacitor  68 . The diode  62 , inductor  64  and capacitors  60 ,  66 , and  74  are tuning elements for tuning to a desired band controlled by the control signal D 4  that sets the center frequency of the desired band. The second DC blocking capacitor  66  is connected to an HP5082-3081 diode  70  functioning as a current-controlled resistor. The diode  70  allows adjustment of the isolation BPF passband amplitude flatness and group delay. The diode  70  is connected to a resistor  72  that is controlled by the control signal D 3  that controls the bandwidth of the bandpass. The resistor  72  and diode  70  are further connected to a third DC blocking capacitor  74  that provides the BPF output  21 . The three series capacitors  60 ,  66 , and  74  provide DC blocking and affect how the passband of the BPF is shaped. Tunable wideband nulls were generated when the diode  70  functioning as a current controlled rectifier (CCR) with a control voltage set to predetermined nonzero values. 
     Referring to FIGS. 1A,  1 B,  2 , and  3 , and more particularly to FIG. 3, the microcontroller routines for detecting, setting the initial reference-path parameters, and adaptively controlling the null depth are performed under program control that start  80  with the reference path effectively disabled using control signals D 3 , D 4 , D 5  and D 6 , without any nulling of the delay signal. The microcontroller has several standard operational routines. The microcontroller is set up to continually coarsely scan the operating bandwidth for undesired signals. Once an undesired signal is detected, a fine-frequency scan is conducted in order to more accurately locate the undesired signal. Next, the microcontroller sets up the reference path parameters according to predefined look up tables. The residual of the nulled signal is the baseband detection signal  38  that is used as a feedback input to a closed loop nulling system. After setting the reference path parameters, the microcontroller  50  then polls the output value of the analog to digital converter  40 . If the residual of the nulled signal is not small enough, the microcontroller adapts the reference path parameters including a reference BPF  20  providing a tuning location, gain tuning value of VGA  26 , and phase delay values of the phase shifter  24  to reduce the value of the residual null signal even further. Once the microcontroller  50  has achieved the best possible null, the controller monitors the results. If the signal drifts in frequency or power, the controller adapts the null to these conditions. Certain defined conditions, such as loss of signal, will cause the microcontroller to return to the search mode. Multiple nulls can be generated if multiple reference signals are obtained through the use of a power divider instead of the current single arm coupler using a plurality of reference paths. The microcontroller  50  can then cycle through setting up each reference path to generate a null at different locations. 
     The microcontroller  50 , that may be an RPC-2300 development system, causes the tunable detection path BFP  30  to scan the input frequency band in coarse steps  82  by using control signal D 2 . At each frequency step, the output signal of the signal detection path BPF  30  is sampled  84  using the converter  40  and the value of the detected signal  38  is read by the microcontroller  50 . The amplitude of the detected signal  38  can be controlled using the control signal D 1  so as to prevent saturation of the diode  34  or so as to amplify small amplitude undesired signals. After the entire frequency band has been scanned, the microcontroller  50  decides whether an undesired signal has been detected. The signals must meet certain criteria in order to be considered undesired signals. A VGA  32  adjusts the input signal level to the square law detection diode  34  in order to increase the range of detectable signals. When an undesired signal has been detected  84 , the signal may be classified by amplitude, frequency, and pulse or CW mode. After an undesired signal has been detected in the coarse scan steps, a fine-frequency scan  86  occurs in order to determine the detected bandwidth and center frequency of the undesired signal. After detecting the undesired signal in the detection path, the microcontroller set the parameters  88  of the reference path to generate a null of the undesired signal. A look up table can be used to cross reference the frequency, amplitude and mode to values of the control signal D 3 , D 4 , D 5  and D 6  for controlling the operation of the reference path. After setting the reference path parameters by controlling signals D 3 , D 4 , D 5  and D 6 , the detection signal  38  is again sampled  90  for any residuals  92  to determine if the nulling has been effective. The detected signal  38  is sampled and control signal D 1  can be varied to measure the amplitude of the detected signal. If the nulling has been effective, then the detection signal is continuously monitored  94  and the microcontroller  50  determines  96  if the nulling has remained effective. The detected power level of the residual of the undesired signal is a measure of the nulling efficiency. When the initial residuals of the detection signal are too high  92  or have subsequently increased due to a slight drift  96 , then the reference path parameters are again finely adjusted  98 , and the detection signal is again sampled. The reduction of the residuals  90 ,  92  and  98 , or the reduction of the drift  90 ,  92 ,  84 ,  96  and  98  are repeated to reduce the residual or to maintain the undesired detection signal within predetermined limits. The power level of the detected signal  38  is in effect a feedback signal for the closed-loop nulling process. The closed loop nulling process continues until the detected signal disappears  96  and the controller returns to coarse scanning  82  searching for another unwanted signal. When the signal disappears, the reference path is turned off using the control signal D 3 , D 4 , D 5 , and D 6  so that the detected signal  38  is not affected by any nulling function in the reference path, so that the scanning  82  reveals unnulled undesired signals. 
     Referring to FIGS. 1A through 5B, and more particularly to FIGS. 4A,  4 B,  5 A, and  5 B, in a narrowband application, an interfering signal is to be cancelled at the center frequency. The phase information is presented as the offset from a 180° difference in the delay path and reference-path phase transfer functions and set by the phase shifter  24 . The delay path has a uniform amplitude transfer function  100 . The reference BPF  20  is set at a desired center frequency  101  where the narrowband amplitude transfer function  102  has a maximum amplitude response. The transfer function defines the shape of the response  102  of the BPF  20  characterized by a Q value. When superimposed in the frequency domain over a signal, the response  102  serves to isolate an interfering signal for cancellation. The current controlled resistor  70  in reference path BPF  20  is set at a high-impedance state, and the Q of the filter is at a maximum. If only such narrowband nulls were to be achieved, then the Q of the reference path BPF should be made as large as possible. The delay path has a uniform phase shift response  104  and matches the narrow band phase transfer function  104  of the BPF  20  only at a certain frequency  105  that corresponds to the selected center frequency  101 . A narrowband null is achieved by matching the amplitude and phase only at this center frequency  101 . Wideband nulls can be generated by flattening and widening the bandpass shape of the reference bandpass filter  20 . The delay path retains a flat amplitude transfer function  108 . However, the reference BPF  20  is controlled by CCR  70  to have a wide band transfer function  110  centered at the center frequency  109  having a maximum amplitude  109  at the center frequency  109 . Biasing the CCR  70  so that its impedance drops to 100 ohm will cause the required reshaping of the reference path BPF  20 . The wideband transfer function defines the shape of the response  110  of the BPF  20  characterized by another Q value. The phase transfer function  112  of the BPF  20  during wideband nulling has a saw tooth phase shift response  104  and is matched  113  only at certain frequencies to the delay path phase transfer function  114  of the BPF  20 . A narrowband null is achieved by matching the amplitudes and phases at only one frequency. In order to generate a wideband null, the BPF amplitude transfer function must be widened, and the group delay of the delay and reference path must match within an offset. A compromise between a perfect amplitude match and a perfect phase match is required. 
     The ideal reference path BPF configuration would be that of a tunable brick-wall filter having a variable bandwidth and group delay that is matched to that of the delay path. An alternative could be to improve the steepness of the null sides of the filter response of the BFP  20  to use a programmable acoustic transversal filter whose phase and amplitude characteristics can be controlled independently over a limited bandwidth. Though the null generation would be improved, the transversal filter would cost more and consume more surface area than does the lumped element BPF  20 . 
     Referring to all of the Figures, and more particularly to FIGS. 6A,  6 B,  7 A, and  7 B, the cancellation system can cancel undesired signals and pass desired signals. There is an assumption that the desired signal and the interfering signal are frequency isolated so that cancellation of the interfering signal is possible without simultaneously canceling the desired signal. In the case of a wideband cancellation, a desired signal  116  and a wideband undesired signal  118  are received as the input signal  10 . After wideband cancellation by isolating, phase shifting and adding the wideband signal back into the input signal  10  using the reference path and directional coupler  14 , to effectively subtract the wideband signal  118  from the input signal, the undesired wideband signal  118  is effectively removed save only an undesired wideband remnant portion  120 . Likewise, in the case of a narrowband cancellation, a desired signal  122  and a narrowband undesired signal  124  are received as the input signal  10 . After narrowband cancellation by isolating, phase shifting and adding the wideband signal to the input signal  10  using the reference path and directional coupler  14 , to effectively subtract the narrowband signal  124  from the input signal  10 , the undesired narrowband signal  124  is effectively removed save only an undesired narrowband remnant portion  126 . 
     The cancellation system samples the interfering signals of the input signal by sampling the detection baseband signal at selected bandwidths. The cancellation system detects an interfering signal and characterizes the interfering signal with respect to frequency, pulse-width length or cw, and amplitude. The baseband detection signal  38  is characterized as to pulse-width or cw, amplitude, and frequency bandwidth and these parameters can be compared against parameter definitions of the undesired signals that are stored in the microcontroller  50  to determine if the detected signal is desired or undesired. Characterization of the baseband detection signal is achieved in the digital control circuit using the pulse width detection circuit  44 , cw detection circuit  46 , threshold comparator  42  and the A/D converter  40 . Once an interfering cw or pulse signal is detected, the microcontroller  50  initiates the cancellation sequence. The polling for interfering signal detection occurs at a predetermined rate, for example, every 0.05 s, the shortest internal programmable delay time allowed by the microcontroller. The microcontroller  50  operates to multiplex the polling of the signal detection from the pulse detection circuit  44  or the cw detection circuit  46 . The pulse width detection circuit  44  rejects cw signals but detects pulse width signals. The pulse width detection circuit  44  detects rising and falling edges of the pulse for determining the timing duration of the pulse. The circuit  44  compares the width of the detected pulse to a preset pulse width, for example, 200 ns. If the detected pulse is longer or shorter than this preset value, then a pulse detection signal is communicated to the microcontroller  50 . Concurrently, the amplitude of the pulse is measured by the A/D converter  40 . The cw detection circuit  46  operates once the detection BPF  30  is set to a specific frequency. The microcontroller . 50  samples the output of the comparator  42  through the cw detection circuit  46  and the selector  48  every 100 ns for 1 ms. The 1 ms time limit is the dwell time that the detection BPF  40  stays at a single frequency. If a cw signal is detected for a greater than a predetermined masking period so as to reject pulse signals, then the cw signal detect condition is set. The masking period and the dwell time are predetermined values stored in the microcontroller  50 . The cw detection circuit  46  provides a cw detection signal to the microcontroller. 
     The microcontroller  50  stores the definition of the undesired signals, manages signal detection, characterizes any undesired signals, outputs control signals to the reference path for phase shifting, AGC setting and bandwidth frequency nulling, monitors the nulling efficiency, and adjusts the reference path to optimize the null depth. The microcontroller  50  first scans the frequency range searching for detected signal in the detection path across the 60 MHz wide operating bandwidth in coarse 2 MHz steps to detect a signal. The desired detection threshold is loaded into the comparator  42  at each frequency location. The microcontroller  60  then monitors the multiplexed outputs of the pulse and cw detection circuitry  44  and  46 . If a detected signal does not meet the undesired signal definition, the controller just repeats the scanning process. During all scanning operations, the signal transmission through the reference path would be disabled by setting the attenuation of the reference-path VGA to a maximum value. 
     When an interfering cw signal whose baseband amplitude exceeds that of the detection threshold is being received in the operating bandwidth during the coarse-scan process, the frequency location, coarse-bandwidth, and amplitude of this undesired signal is determined. After a scan of the entire operating bandwidth had been completed, the center frequency location of the selected largest amplitude signal is determined. The microcontroller  50  then starts a fine frequency scan within ±5 MHz around the center frequency in steps of 0.4 MHz in order to determine the magnitude and frequency bandwidth of the interfering signal around the center frequency. This bandwidth determination is used to set the controller to narrowband or wideband mode. During the fine frequency scans, the controller adaptively adjusts the AGC setting in the detection path to avoid saturating the detection diode  34 . In this manner, an automatic gain control process is used to adjust the 20 dB dynamic range of the detection diode  34  to bracket the amplitude of the detected interfering signal. 
     The controller contains a predetermined lookup table of the parameter values of the reference path phase shift, AGC setting and bandwidth tuning voltages of the reference path. The coarse table consists of only a few frequency points. An interpolation routine uses the center frequency to determine the required reference path parameter values. The interpolated values are then output to the reference path components using the control signal D 3 , D 4 , D 5  and D 6 . This adjusted version of the reference signal is then vector summed in the directional coupler  14 . In order to detect the reduced amplitude of the interfering signal, the AGC attenuation in the detection path is decreased by an assumed set value of approximately 20 dB. Thus, any remnant of the interfering signal can be measured. 
     The controller then monitors the magnitude of the remnant of the interfering RF signal, and adjusts the reference path parameters to minimize the value of the remnant residual interfering signal. The values of the reference path components are separately adjusted ±10% around the previously determined interpolated values. A simple minimum value search routine is used to determine the optimum set of reference path parameter settings that minimize the value of the residual signal. If the optimization process causes the magnitude of the residual signal to drop below the detection threshold, the AGC attenuation setting of the detection-path VGA is decreased so that the residual signal can again be detected. At the end of the optimization process, the controller monitors the final residual value of the interfering signal. Several conditions are tested to determine whether the interfering signal is drifting in amplitude or frequency. 
     The microcontroller  50  can employ various algorithms for the detection and cancellation of undesired signal. Square law detection in combination with threshold level determines the presence of a signal. In the preferred form, a detected signal is determined to be a desired or undesired pulse signal, relative to the preset pulse duration. If desired, the signal is determined to be a desired or undesired cw signal. CW or pulse mode detection, square law amplitude detection and frequency scanning detection are used to characterize a detected signal. A lookup table can be used to identify the required reference path control signal D 3 , D 4 , D 5  and D 6  values. The frequency location of the undesired signal is cross referenced in a lookup table to determine reference path parameters for controlling the reference path to canceling the undesired signals from the composite input signal consisting of both desired and undesired signals. The lookup table may be, by way of example for narrowband detection, a table having a plurality of scanned detection center frequencies F 1 , F 2 , F 3  through Fn cross referenced to respective VGA amplitudes A 1 , A 2 , A 3  through An, phase θ 1 , θ 2 , θ 3 , through θn, and BPF center frequencies f 1 , f 2 , f 3 , through fn. When, for example, a signal is detected as a narrowband signal at a center frequency of F 2 , the reference path is controlled to have a center frequency of f 2 , a phase shift of θ 2 , and an amplitude of A 2 , but over a predetermined narrowband of operation. The lookup table may further include, by way of example for wideband detection, a lookup table having a plurality of scanned detection center frequencies F 1 , F 2 , F 3 , through Fn over a predetermined wideband and cross referenced to respective VGA amplitudes A 1 , A 2 , A 3 , through An, phases θ 1 , θ 2 , θ 3 , through θn, and BPF center frequencies f 1 , f 2 , f 3 , through fn, and current control resistor values R 1 , R 2 , R 3 , and RN, for providing bandwidths b 1 , b 2 , b 3 , and bN, respectively. When, for example, a signal is detected over a wideband by having substantial amplitudinal component over the wideband, but centered at a center frequency of F 2 , the reference path is controlled to have a center frequency of f 2 , a phase shift of θ 2 , an amplitude of A 2 , a current controlled resistor value R 2 , for providing a bandwidth b 2 , of the reference BPF  20 . In this manner, the microcontroller  50  can detect the presence of signal, characterize the detected signal, and if unwanted, automatically control the reference path to cancel the unwanted signal. 
     The present invention is characterized by a detection path for detecting the presence of desired and undesired signals. When an undesired signal is detected, a reference path is controlled to isolate the undesired signal and subtract it from the input signal so that only the desire signal survives without apriori information of the undesired signal and without the use of beam nulling or external undesired signal samples. Those skilled in the art can make enhancements, improvements, and modifications to the invention, and these enhancements, improvements, and modifications may nonetheless fall within the spirit and scope of the following claims.