Abstract:
An active rectifier including an anode terminal, a cathode terminal, a MOSFET operable to allow current flow from the anode to the cathode, a voltage sensing circuit operable to sense the voltage between the anode and cathode terminals and a gate drive circuit responsive to the voltage sensing circuit and operable to drive the gate of the MOSFET such that the MOSFET conducts when the voltage at the cathode terminal is less than that at the anode terminal and such that the conduction occurs substantially in the linear operating region of the MOSFET.

Description:
FIELD OF INVENTION 
     This invention relates to an active rectifier. 
     BACKGROUND ART 
     Conventional rectifier circuits use either bipolar or Schottky diodes, both of which suffer from deficiencies. 
     Firstly, bipolar diodes have a higher forward voltage drop than Schottky diodes, so tending to introduce unacceptable power losses in high current circuits. In addition, reverse recovery currents can be large if the diode is switched from full conduction to blocking in a low impedance circuit. These currents can increase circuit stresses and power losses, as well as contributing to EMI (electromagnetic interference) from the host equipment. 
     Secondly, Schottky diodes, while possessing the advantage of lower forward voltage drops than comparable bipolar types, tend to show a more rapid rise of leakage current with temperature. This can lead to excessive power dissipation if reverse voltages are high. 
     It has been recognised that a power MOSFET (metal oxide semiconductor field effect transistor) possesses good characteristics for a rectifier. The forward voltage drop in ‘gate-on’ mode can be very low and there is very little reverse recovery current at low frequencies of operation if the drain/body diode of the MOSFET does not carry the forward current. Also, high leakage currents do not occur and most MOSFETs can avalanche safely when subjected to overvoltage transients. 
     A number of circuits exist in which MOSFETs are used as active rectifiers, but these suffer from problems of control and timing. For example, if the active rectifier drive in a switched mode power supply is taken from the overall control circuit, which is often on the primary side of an isolation transformer, some means must be found to drive the active rectifier on the secondary side. This might involve the use of a second transformer. Because of different leakage inductance effects in the two transformers, this arrangement can give rise to imperfect timing between the main transformer output and the active rectifier drive. This causes power losses. 
     An alternative is to take the active rectifier drive from an overwind on the main transformer. However, this complicates the design of the transformer and is still not as good as using the actual secondary output since leakage inductance can again affect timing. 
     Other solutions offer secondary-side control and are based on switching the MOSFET by means of a comparator circuit. An example of a comparator based rectifier is given on page 237 of Electronics World for March 1999. Although the circuit of the present invention has some similarities to this approach, the design differs at least in that the control circuit and MOSFET operate in the linear mode. 
     This major difference reduces the problems caused by operating devices in the saturated mode, which saturation increases the time taken to change from one state to the other. The result of this time increase in switching the MOSFET OFF gives rise to an effect similar to reverse recovery current in a conventional diode. 
     SUMMARY OF THE INVENTION 
     In accordance with the invention there is provided an active rectifier comprising an anode terminal, a cathode terminal, a MOSFET operable to allow current flow from the anode to the cathode, a voltage sensing circuit operable to sense the voltage between the anode and cathode terminals and a gate drive circuit responsive to the voltage sensing circuit and operable to drive the gate of the MOSFET such that the MOSFET conducts when the voltage at the cathode terminal is less than that at the anode terminal and such that the conduction occurs substantially in the linear operating region of the MOSFET. 
     In a second aspect, the invention provides a rectifier circuit including the active rectifier of the first aspect. 
     In a third aspect, the invention provides a regulator circuit including the active rectifier of the first aspect. 
     In a fourth aspect, the invention provides a converter circuit including the active rectifier of the first aspect. 
     Embodiments of active rectifiers in accordance with the invention and applications of the active rectifier will now be described by way of example with reference to the drawings. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a schematic circuit diagram of the active rectifier connected in a simple inductive circuit. 
     FIG. 2 is an alternative embodiment of the circuit of FIG. 1 with the compensation diode formed by a bipolar transistor in a “super-diode” configuration. 
     FIG. 3 is a further embodiment of the circuit shown in FIG. 2, including a frequency compensation network for circuit stability. 
     FIG. 4 is a further embodiment of the circuit shown in FIG. 3, including a circuit to increase the speed of turn ON of the rectifier by transiently overdriving the MOSFET. 
     FIG. 5 is a schematic symbol for the rectifier circuit. 
     FIG. 6 is a schematic diagram of a bi-phase rectifier circuit incorporating the active rectifier. 
     FIG. 7 is a schematic circuit diagram of a full bridge rectifier circuit incorporating the active rectifier. 
     FIG. 8 is a schematic circuit diagram of a buck regulator circuit incorporating the active rectifier. 
     FIG. 9 is a schematic circuit diagram of a CUK converter incorporating the active rectifier. 
     FIG. 10 is a schematic circuit diagram of a transformer-coupled flyback converter incorporating the active rectifier. 
     FIG. 11 is a schematic circuit diagram of a boost converter incorporating the active rectifier. 
     FIG. 12 is a schematic circuit diagram of a flyback converter circuit incorporating the active rectifier. 
     FIG. 13 is a schematic circuit diagram of a forward converter circuit incorporating the active rectifier. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     With reference to FIG. 1, an active rectifier consists of a field effect transistor TR 4  together with a drive part (shown within a dotted outline) which could be manufactured as an integrated circuit. For the purposes of illustrating the operation of the active rectifier, FIG. 1 also shows a power supply PSU, a switch S 1 , a capacitor C 1  and a load inductance L. 
     In addition to the field effect transistor TR 4 , the drive part consists of a conventional gate drive transistor pair TR 2  and TR 3  with a current pull-up CS 1  (such as a current diode), and a voltage sensing circuit consisting of R 1 , TR 1 , D 1  and D 2 . Although an auxiliary supply Vcc is shown as being required for the circuit to operate, such a supply can be obtained from the reverse voltage across the rectifier. Suggestions for its derivation are shown in the Figures. 
     In low frequency circuits it is possible to replace CS 1  by a resistor. D 1  provides temperature compensation for the base/emitter voltage drop of TR 1 . It also eliminates the effect of that voltage drop on the operation of the circuit. C 1  ensures a low impedance source for gate drive transistor TR 2 . 
     With S 1  closed, the voltage across the inductor L causes the current flowing from the power supply to increase at a constant rate. Because the voltage across TR 4  equals that of the supply, D 1  and D 2  are both reversed biased, thus allowing all the current flowing in R 1  to flow into the base of TR 1 . This maintains TR 1  in an ‘ON’ state, which holds the gate of TR 4  at zero. TR 4  is therefore ‘OFF’. 
     When S 1  opens, the voltage on the drain of TR 4  falls rapidly until it reaches the point below 0V where D 1  and D 2  begin to conduct in the forward direction. Current is then diverted from the base of TR 1 , resulting in TR 1  beginning to switch ‘OFF’ and TR 4  beginning to turn ‘ON’. With TR 4  in conduction, inductor current flows upwards through it, from the anode of the rectifier to the cathode. TR 4  does not saturate because the drive part receives feedback via D 1  and D 2 . The ‘ON’ voltage across TR 4  is thereby adjusted by TR 1 , to equal the forward voltage drop of D 2 . This diode provides the reference voltage for the active rectifier feedback circuit. For good circuit efficiency, the voltage across D 2  is made low by an appropriate choice of R 1 . 
     The significant advantage of feedback operation of the overall circuit is that no device operates in a saturated condition, which is an important factor in the ability of the circuit to move rapidly from a conducting to a non-conducting state. In this embodiment, the critical element is TR 1 , which varies the level of drive to the gate of TR 4  to satisfy the voltage balance requirement described at the end of the previous paragraph. Because it does not saturate when TR 4  is conducting, the problems of storage time, due to base overdrive in this device, never arise. 
     It is important that the drain/source parasitic diode of TR 4  does not conduct immediately prior to circuit turn-off, otherwise “reverse-recovery” currents can be generated. This is especially true for high-frequency operation. 
     In practice the rectifier system cannot turn on instantaneously because the gate-drive voltage risetime is limited by drive current and MOSFET gate capacitance effects. Drive current itself is limited by the value of CS 1  and the current gain of TR 2 . The current from CS 1  is affected by local parasitic capacitors. The most significant of these is the collector base capacitance of TR 1 . When S 1  opens, the initial clamping of the negative-going voltage on the drain of TR 4  is accomplished by its drain/source parasitic diode. Conduction time in this mode is determined by the speed at which the drive circuit can establish an appropriate level of gate drive to give linear operation. With the appropriate choice of components the parasitic diode conduction period can be made a small fraction of the total conduction time, so maintaining the overall efficiency of the circuit. 
     The turn-off of the circuit will now be described. 
     When S 1  closes, current flowing in TR 4  is diverted into the driving power supply, which causes the channel resistive voltage drop of TR 4  to fall below that which TR 1  is attempting to maintain. As a consequence of the feedback mechanism described above, the drive to TR 4  reduces to zero because its channel voltage drop no longer exists. TR 4  is now switched ‘OFF’. 
     In practice, current will flow into TR 4  as its voltage rises because drain/source and drain/gate parasitic capacitors need to be charged. Of these, the latter is more significant because the drive transistor TR 3  needs to be able to sink the charging current. In turn, this requires that TR 1  can sink the base drive current of TR 3 . Failure to hold the gate of TR 4  at 0V during the voltage-rise period increases its switching loss. Both of these parasitic capacitors, and the finite operating time of the driver, give rise to an effect similar to reverse-recovery current in a bipolar diode. 
     One factor in this “reverse-recovery current” is the finite time that it takes for the gate drive to reduce to zero when load current is diverted out of TR 4 . If this diversion is very rapid, TR 4  may still be in an ON state as the voltage across it is attempting to rise. This, effectively, transient short circuit caused during the turn-off period can give rise to a large pulse of current. 
     Also, it is important that the temperature coefficients of TR 1  (Vbe) and D 1  are matched, otherwise it is possible for the effective voltage reference provided by D 2  (to ensure linear operation of the circuit) to be enhanced by the voltage differential between D 1  and the Vbe of TR 1  and to be above the forward voltage drop of the parasitic diode of TR 4 . In this condition, the effective reverse-recovery current can increase due to conduction in the parasitic diode, giving rise to higher circuit losses. A solution is to use a transistor of the same type as TR 1  (preferably thermally coupled to it) in a “super-diode” configuration to ensure good temperature tracking. 
     FIG. 2 shows a circuit incorporating this modification where D 1  has been replaced. by super-diode TR 5 . 
     As will be seen, a small inductor L 1  has also been inserted in series with the drain of TR 4  L 1  provides control of reverse recovery current. It is not part of the basic active rectifier circuit but is used for efficient performance at high frequency. L 1  should be a non-linear inductor since it is required to saturate rapidly when drain current flows. It may, for example, be a small bead made from low-loss ferrite, or a bead made from ‘square loop’ amorphous strip. The purpose of this component is to provide a buffer between TR 4  and the source supply when S 1  closes. 
     With S 1  open, current flows upwards through TR 4  and L 1  is saturated. This is as described in connection with FIG.  1 . When S 1  closes, current transfers instantaneously into the source power supply but no “reverse recovery” current flows in TR 4  because of the inductance of L 1 . Since there is now no ‘forward’ voltage drop across TR 4 , the driver stage starts to switch the gate to OFF. The drive for this part of the operation is supplied via R 1  into the base of TR 1 . 
     L 1  must provide sufficient hold-off (the ability to withstand an applied voltage for a given period of time) when S 1  closes, otherwise the drive to TR 4  may not be fully ‘OFF’ before it saturates. Ideally, the core should just be on the point of saturation (fully reset) when the voltage across TR 4  has reached that of the source supply. This means that reset is dependant, not only on delay time in the driver, but on the time it takes to charge the parasitic drain capacitors of TR 4 . For this reason, a square-loop material is more suitable for the core because it provides the maximum amount of hold-off for a given size. It should be noted that the flux swing in the core of L 1  is very wide and precautions may need to be taken to ensure that it does not overheat when used in high frequency circuits. 
     FIG. 3 shows the addition of resistor R 2 , resistor R 3  and capacitor C 2 , which are used to ensure stability because the whole circuit operates in a linear feedback mode when TR 4  is conducting as a rectifier. 
     FIG. 4 shows the addition of transistor TR 6 , resistor R 4 , resistor R 5  and capacitor C 3 . These components provide regenerative feedback to give gate overdrive for TR 4  at the instant that rectification commences. TR 6  is a high-speed switching transistor and the values of R 4 , R 5  and C 3  are chosen to control the period of overdrive to the required value. 
     The operation of the circuit is such that the base of TR 1  is shorted to 0V by TR 6  when the drive voltage to TR 4  begins to rise. The rate-of-rise of voltage is partly controlled by the charging of the parasitic collector/base capacitor of TR 1 . By preventing this charging current from flowing into the base, the rate-of-rise of collector voltage, and hence TR 4  gate voltage, can be increased. This gives a slight improvement in rectifier efficiency because conduction time in the body diode of TR 4  is reduced because of the faster transition to controlled conduction. 
     If the overdrive time is chosen correctly, the circuit will make a transition from saturated to linear operation shortly before the active rectifier is required to turn OFF. In this way, the advantages of both linear and saturated operation can be realised. Saturated operation produces the highest efficiency; linear operation at the instant of turn OFF raises the efficiency by reducing the equivalent reverse recovery current. 
     FIG. 5 shows a symbol which is used in the subsequent Figures to denote the circuit of FIG.  4 . It will be noted that a zener diode D 3  is connected between the auxiliary power supply Vcc and the anode. The purpose of this diode is to ensure that the gate drive of TR 4  never exceeds a safe value irrespective of the source of the auxiliary supply. Thus each of the circuits described below also incorporate such a zener diode. 
     Furthermore, all of the circuits described below show a resistive feed from the most positive part of the power circuit to Vz. However, in most cases where the anode of the rectifier circuit is referenced to 0V, a local auxiliary supply may be available. A diode has also been included in the Vz feed in the circuits described below. This has been included to prevent the local Vz decoupling capacitor from discharging through the feed resistor when the rectifier circuit is in full conduction. 
     FIG. 6 shows a bi-phase rectifier circuit incorporating two of the active rectifiers described above. 
     FIG. 7 shows a full bridge rectifier circuit including four of the active rectifiers described above. It should be noted that although this Figure shows a single phase rectifier circuit, the addition of a third arm would allow three phase rectification. The active rectifier circuit could also be operated in a star or delta configured circuit and in a polyphase system. It is to be noted that the circuits of FIGS. 6 and 7 could operate on a sinusoidal input. The active rectifier is particularly advantageous in such applications since additional drive windings are not required because the circuits are self-commutating. 
     FIG. 8 shows a buck-regulator circuit incorporating the active rectifier circuit described above. 
     FIG. 9 shows a CUK converter incorporating an active rectifier as described above. 
     FIG. 10 shows a transformer-coupled flyback converter incorporating an active rectifier as described above. 
     FIG. 11 shows a boost converter circuit incorporating an active rectifier as described above. 
     FIG. 12 shows a flyback converter circuit incorporating an active rectifier as described above. 
     FIG. 13 shows a forward converter circuit incorporating the active rectifier described above. For the circuit of FIG. 13, the series, saturable inductor is not required for the active rectifier circuit because the reverse voltage is driven by the magnetising current of T 1  when TR 1  turns off. As such, the rate of rise of reverse voltage is controlled by a current source which, by definition, cannot give rise to high recovery currents in the active rectifier circuit. 
     Since the components of active rectifier described above operate in their unsaturated modes and in particular, the MOSFET TR 4  operates in its linear or resistive region, switching times are fast and losses are small. Thus, this circuit has applications in fields, such as the aerospace industry where circuit efficiency is important. 
     It will be appreciated that although MOSFET TR 4  has been shown and described as an n-channel MOSFET, it is, however, possible to operate the circuit using a p-channel MOSFET. In that case, the npn bipolar transistors are swapped for pnp bipolar transistors and vice versa and the polarities of the supplies and diodes are reversed. 
     It will also be appreciated that the Schottky diode could be replaced with another diode type having a low forward voltage drop.