Abstract:
An apparatus for transferring electromagnetic energy intermediate a host device and a medium or free space adjacent to the apparatus in an impulse radio system includes: (a) an energy guiding means for guiding the electromagnetic energy; the energy guiding means is connected with the host device; (b) an electromagnetic energy channeling structure effecting the transferring and including a plurality of gap interfaces; and (c) a transition means for coupling the energy guiding means with at least one gap interface of the plurality of gap interfaces. The transition means conveys the electromagnetic energy intermediate the energy guiding means and the at least one gap interface. The at least one gap interface intersects the transition means in a substantially continuous curve in selected planes intersecting the gap interface and the transition means.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates generally to electromagnetic energy radiation and reception, and especially relates to electromagnetic energy radiation and reception effected using impulse radio energy. Still more particularly the present invention provides an antenna suited for broadband energy radiation and reception, and particularly well suited for broadband energy radiation and reception employing impulse radio energy. 
     2. Related Art 
     Recent advances in communications technology have enabled an emerging revolutionary ultra wideband technology (UWB) called impulse radio communications systems (hereinafter called impulse radio). 
     Impulse radio was first fully described in a series of patents, including U.S. Pat. No. 4,641,317 (issued Feb. 3, 1987), U.S. Pat. No. 4,813,057 (issued Mar. 14, 1989), U.S. Pat. No. 4,979,186 (issued Dec. 18, 1990) and U.S. Pat. No. 5,363,108 (issued Nov. 8, 1994) to Larry W. Fullerton. A second generation of impulse radio patents include U.S. Pat. No. 5,677,927 (issued Oct. 14, 1997) to Fullerton et al; and U.S. Pat. No. 5,687,169 (issued Nov. 11, 1997) and U.S. Pat. No. 5,832,035 (issued Nov. 3, 1998) to Fullerton. These patent documents are incorporated herein by reference. 
     Uses of impulse radio systems are described in U.S. patent application Ser. No. 09/332,502, entitled, “System and Method for Intrusion Detection Using a Time Domain Radar Array,” and U.S. patent application Ser. No. 09/332,503, entitled, “Wide Area Time Domain Radar Array,” both filed Jun. 14, 1999, both of which are assigned to the assignee of the present invention, and both of which are incorporated herein by reference. 
     Basic impulse radio transmitters emit short pulses approaching a Gaussian monocycle with tightly controlled pulse-to-pulse intervals. Impulse radio systems typically use pulse position modulation, which is a form of time modulation where the value of each instantaneous sample of a modulating signal is caused to modulate the position of a pulse in time. 
     For impulse radio communications, the pulse-to-pulse interval is varied on a pulse-by-pulse basis by two components: an information component and a pseudo-random code component. Unlike direct sequence spread spectrum systems, the pseudo-random code for impulse radio communications is not necessary for energy spreading because the monocycle pulses themselves have an inherently wide bandwidth. Instead, the pseudo-random code of an impulse radio system is used for channelization, energy smoothing in the frequency domain and for interference suppression. 
     Generally speaking, an impulse radio receiver is a direct conversion receiver with a cross correlator front end. The front end coherently converts an electromagnetic pulse train of monocycle pulses to a baseband signal in a single stage. The data rate of the impulse radio transmission is typically a fraction of the periodic timing signal used as a time base. Because each data bit modulates the time position of many pulses of the periodic timing signal, this yields a modulated, coded timing signal that comprises a train of identically shaped pulses for each single data bit. The impulse radio receiver integrates multiple pulses to recover the transmitted information. 
     In a multi-user environment, impulse radio depends, in part, on processing gain to achieve rejection of unwanted signals. Because of the extremely high processing gain achievable with impulse radio, much higher dynamic ranges are possible than are commonly achieved with other spread spectrum methods, some of which must use power control in order to have a viable system. Further, if power is kept to a minimum in an impulse radio system, this will allow closer operation in co-site or nearly co-site situations where two impulse radios must operate concurrently, or where an impulse radio and a narrow band radio must operate close by one another and share the same band. 
     Many applications for impulse radio technology, including communication applications, position determination applications, locating (e.g., radar) applications and other applications require lightweight, compact broadband antennas with omni-directional transmit/receive characteristics. As with any antenna, impedance matching to feed elements is necessary for efficient operation. Moreover, in the case of impulse radio technology applications, the antenna must not be subject to ringing in response to application of pulses—either in a transmit mode or in a receive mode. 
     Current antenna technology offers several undesirable alternatives to one interested in a small, well-matched, efficient, omni-directional ultra wideband (UWB) short pulse antenna: (1) a self-similar antenna (e.g., a bow tie antenna) that tends to be large and frequency dispersive; (2) a TEM horn antenna that tends to be bulky and highly directive; or (3) a resistively loaded antenna that will necessarily be lossy and inefficient. Existing spheroidal antennas like the volcano smoke antenna ( FIG. 10 ) are difficult to manufacture. Existing UWB antennas like the biconical antenna are relatively large and, despite their stable impedance, are not well matched to 50Ω. 
     Kraus (John D. Kraus,  Antennas,  2 nd  edition; New York: McGraw Hill, 1988) briefly mentions a “double dish” antenna comprised of a pair of hemispherical dishes connected in tandem to form a dipole with planar elements facing away from each other. (Kraus; p. 63) The “double dish” configuration is presented as a step in evolving an antenna configuration from Kraus&#39;s “volcano smoke” antenna ( FIG. 10 ) to a stub (monopole) antenna. Kraus&#39; “double dish” antenna does not meet the performance criteria recognized herein as necessary for optimal performance in an impulse radio application. The sharp discontinuities in transitioning from a smooth curve to a substantially planar outwardly facing dish surface creates undesirable internal reflections in the “double dish” antenna. 
     The current art regarding ultra wideband (UWB) antennas teaches using element antennas such as monopoles, dipoles, conical antennas and bow-tie antennas for ultra wideband systems. However, they are generally characterized by low directivity and relatively limited bandwidth unless either end loading or distributed loading techniques are employed, in which case bandwidth is increased at the expense of radiation efficiency. 
     Conventional antennas are designed to radiate only over the relatively narrow range of frequencies used in conventional narrow band systems. Such narrow band systems may, for example, employ fractional bandwidths no more than about 25%. If an impulse signal, such as a signal of the sort employed for impulse radio purposes, is fed to such a narrow band antenna, the antenna tends to ring. Ringing severely distorts signal pulses and spreads them out in time. Impulse radio signals are preferably modulated by pulse timing, so such distortion of pulses is not desirable. 
     Broadband antennas are advantageous for many purposes, including their use with impulse radio systems. Conventional design in broadband antennas follows a commonly accepted principle that the impedance and pattern properties of an antenna will be frequency independent if the antenna shape is specified only in terms of angles. That is to say, a self-similar or self-complimentary antenna will be a broadband antenna. This principle explains known broadband antennas like biconical and bow tie antennas, but also applies to other broadband antennas like log periodic, log spiral, and conical spiral antennas. 
     All such prior art antennas rely on a variation of scale to achieve their broadband performance. A “smaller” scale section of the antenna radiates higher frequency components while a “larger” scale section of the antenna radiates lower frequency components. Because the radiation centers change location as a function of frequency, these antennas are inherently frequency dispersive; they radiate different frequency components from different parts of the antenna, resulting in a distorted impulse signal. 
     Throughout this description, it should be kept in mind that discussions relating to transmitting or transmissions apply with equal veracity to reception of electromagnetic energy or signals. In order to avoid prolixity, the present description will focus primarily on transmission characteristics of antennas, with the proviso that it is understood that reception of energy or signals is also inherently described. 
     A biconical antenna is a classic example of a prior art broadband antenna with an omni-directional pattern. A typical biconical antenna with a 60° half angle will have a 100Ω input with a voltage standing wave ratio (VSWR) of &lt;2:1 over a 6:1 bandwidth. A significant drawback with such a biconical antenna is that such an antenna is typically implemented with a diameter equal to the wavelength at the lower frequency limit (λ I, ) thus requiring that the antenna be 0.577λ L  in height. Because of similar design limitations, a typical monocone antenna will not provide a good match if it is much less than 0.22λ L  in diameter. In any event, a monocone antenna does not have very stable performance over a broad band. Antennas as large as the above described typical conical antennas (biconical and monoconical) often have difficulty radiating (i.e., transmitting) pulses without dispersion. In addition, such large antennas are difficult to fit into a small portable or hand held devices. 
     TEM horn antennas often suffer from frequency dispersion as well. Furthermore, a horn antenna is inherently a large structure, often several wavelengths in dimension. A horn antenna may be made smaller by dielectric loading, but such loading adds weight which is often undesirable. Further, a horn antenna is a directive antenna and cannot provide the omni-directional coverage required for many portable or mobile applications. 
     A TEM feed may be combined with a parabolic dish to create a ribbed horn “impulse radiating antenna” (IRA). Such antennas can have bandwidths on the order of a couple of decades, and very high gain, but their large size and high directivity make them inappropriate for portable or mobile use. 
     A “dish” antenna consisting of the rounded sides of two spherical hemispheres being driven against one another is a known antenna structure (e.g., Kraus&#39; “double dish” antenna), but it is not known to be used for impulse radio broadband applications. Another prior art attempt to provide a spheroidal antenna is a “volcano smoke” antenna (see, Kraus; p. 63). The tapered feed of this antenna provides excellent matching, and the antenna does radiate omni-directionally, but the gradual transition required to yield such beneficial operating parameters makes the antenna bulky and difficult to manufacture. 
     Because spherical antennas must be fed by a radial waveguide, they often exhibit poor matching characteristics unless an elaborate and difficult-to-manufacture impedance matching structure is used. An impedance matching structure also tends to further impair antenna performance by making the antenna more likely to ring. It is very difficult to construct a feed that maintains a constant matched impedance over a broad bandwidth, something essential to an ultra wideband (UWB) antenna. It is a commonly accepted design criteria in electromagnetic applications, and especially in radio communication applications, that an antenna should match a 50Ω impedance feed providing signals to (or receiving signals from) the antenna. Some video applications require matching a 75Ω impedance feed. 
     Another prior known antenna structure drives a hemispherical antenna against a ground plane. Attempts by the inventor to employ such an antenna structure for broadband impulse radio resulted in an unacceptably large impedance mismatch. 
     Circular disc (planar) monopole antennas and elliptical disc (planar) monopole antennas have been evaluated to determine their respective bandwidths. (Agrawall, Kumar and Ray; “Wide-Band Planar Monopole Antennas”; IEEE Transactions on Antennas and Propagation, February 1998.) However, no regard was given to the suitability of such antennas for impulse radio applications. No regard was given to dispersion, ringing or phase performance of signals employing such circular disc antennas or elliptical disc antennas for impulse radio communication. 
     Resistive loading is an alternate technique commonly employed to achieve impedance matching in broadband antennas. Resistive loading succeeds in reducing reflection, but at the cost of throwing away typically around half the power that may be transmitted by an antenna. Such a design trade-off has become accepted in design approaches in prior art antennas. It has been generally believed that resistive loading must be employed for a small broadband antenna in order to achieve good impedance matching. Non-resistively loaded small ultra wideband antennas are known, but they tend to have poor impedance matching and high voltage standing wave ratios (VSWR&#39;s). A lower value for VSWR is a better value; the optimum value of VSWR is 1:1. The prior art teaches that resistive loading must be used in an element antenna in order to achieve wide bandwidth. It is commonly believed that high radiation efficiency and high bandwidth are mutually exclusive. 
     For a small hand held or portable system, it is desirable to have a well matched, efficient, physically small, UWB antenna that radiates non-dispersively and omni-directionally. It is particularly advantageous for an antenna to be easily made in large volumes with reliable repeatable quality. Not only are such antennas unknown to the present art, in fact, the current teaching is that such antennas are not physically realizable. 
     There is a need for a broadband antenna that is compact, efficiently matched to a feed structure and radiates omni-directionally. 
     In particular, there is a need for a broadband antenna that operates without ringing in response to application of a pulse signal. 
     SUMMARY OF THE INVENTION 
     An apparatus for transferring electromagnetic energy intermediate a host device and a medium or free space adjacent to the apparatus in an impulse radio system comprises: (a) an energy guiding means for guiding the electromagnetic energy; the energy guiding means is connected with the host device; (b) an electromagnetic energy channeling structure effecting the transferring and including a plurality of gap interfaces; and (c) a transition means for coupling the energy guiding means with at least one gap interface of the plurality of gap interfaces. The transition means conveys the electromagnetic energy intermediate the energy guiding means and the at least one gap interface. The at least one gap interface intersects the transition means in a substantially continuous curve in selected planes intersecting the at least one gap interface and the transition means. 
     An energy guiding means is preferably embodied in a structure that conveys electromagnetic energy. Examples of an energy guiding means include, by way of illustration and not by way of limitation, coaxial cable, stripline, microstrip, twin lead, twisted pair fiber optic cable, wave guide or other transmission line, or a connector or coupler that enables connection to a transmission line. 
     An energy channeling structure is preferably embodied in a structure that couples electromagnetic energy between an apparatus and an adjacent free space or medium. Examples of a channeling structure include, by way of illustration and not by way of limitation, radiating elements, receiving elements, reflectors, directors and horns. 
     A transition means is preferably embodied in a structure that receives radio frequency (RF) energy, transmits RF energy or receives and transmits RF energy. The terms “feed” or “feed region” are sometimes used to refer to a transition means. 
     A host radio is a RF device that receives RF energy, transmits RF energy or receives and transmits RF energy. An antenna may be integrally included with or within a host radio or that antenna may be situated remotely from the host radio at an arbitrary distance yet coupled with the host radio, such as by using an energy guiding means. The term “host radio” does not per se indicate any particular relation between a radio and an associated antenna. In particular, the term “host radio” does not preclude an antenna remotely located from a radio or standing alone with respect to a radio. 
     The inventor has discovered that the preferred construction of the electromagnetic energy channeling structure is in a spheroidal or ovoidal shape. The terms “spheroidal” or “ellipsoidal” are employed herein to indicate a three-dimensional element having a generally smoothly curved shape. In its most preferred embodiment, a “spheroidal” or “ellipsoidal” element presents planar sections oriented substantially symmetrically about at least one axis. Thus, a preferred embodiment of a “spheroidal” or “ellipsoidal” element presents a substantially continuously curved intersection with a gap interface in a planar section in an antenna. The curved intersection is bounded by termination loci substantially at the limit of or outside a feed region. The boundary of the “spheroidal” or “ellipsoidal” element departs from each termination locus in a substantially smooth, continuous curve to the other termination locus. The curve is substantially smooth and continuous in dimensions that are significant with regard to the wavelengths with which the element is employed. 
     The terms “ovoidal” or “elliptical” are employed herein to indicate a substantially two-dimensional, planar element having a generally smoothly curved shape. In its most preferred embodiment, an “ovoidal” or “elliptical” element is oriented substantially symmetrically about at least one axis. Thus, a preferred embodiment of an “ovoidal” or “elliptical” element presents a substantially continuously curved intersection with a gap interface in a plane in an antenna. The curved intersection is bounded by termination loci substantially at the limit of or outside a feed region. The boundary of the “ovoidal” or “elliptical” element departs from each termination locus in a substantially smooth, continuous curve to the other termination locus. The curve is substantially smooth and continuous in dimensions that are significant with regard to the wavelengths with which the element is employed. 
     It is therefore an object of the present invention to provide an apparatus for transferring electromagnetic energy intermediate a host device and a medium adjacent to the apparatus that is efficient in operation and easy to manufacture in production level quantities. 
     It is a further object of the present invention to provide an apparatus for transferring electromagnetic energy intermediate a host device and a medium adjacent to the apparatus that is compact and is matched to a feed structure. 
     It is yet a further object of the present invention to provide an apparatus for transferring electromagnetic energy intermediate a host device and a medium adjacent to the apparatus that radiates omni-directionally. 
     It is a still further object of the present invention to provide an apparatus for transferring electromagnetic energy intermediate a host device and a medium adjacent to the apparatus that operates without ringing in response to application of a pulse signal. 
     Further objects and features of the present invention will be apparent from the following specification and claims when considered in connection with the accompanying drawings, in which like elements are labeled using like reference numerals in the various figures, illustrating the preferred embodiments of the invention. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1A  illustrates a representative Gaussian Monocycle waveform in the time domain. 
         FIG. 1B  illustrates the frequency domain amplitude of the Gaussian Monocycle of FIG.  1 A. 
         FIG. 2A  illustrates a pulse train comprising pulses as in FIG.  1 A. 
         FIG. 2B  illustrates the frequency domain amplitude of the waveform of FIG.  2 A. 
         FIG. 3  illustrates the frequency domain amplitude of a sequence of time coded pulses. 
         FIG. 4  illustrates a typical received signal and interference signal. 
         FIG. 5A  illustrates a typical geometrical configuration giving rise to multipath received signals. 
         FIG. 5B  illustrates exemplary multipath signals in the time domain. 
         FIGS. 5C-5E  illustrate a signal plot of various multipath environments. 
         FIGS. 5F  illustrates the Rayleigh fading curve associated with non-impulse radio transmissions in a multipath environment. 
         FIG. 5G  illustrates a plurality of multipaths with a plurality of reflectors from a transmitter to a receiver. 
         FIG. 5H  graphically represents signal strength as volts vs. time in a direct path and multipath environment. 
         FIG. 6  illustrates a representative impulse radio transmitter functional diagram. 
         FIG. 7  illustrates a representative impulse radio receiver functional diagram. 
         FIG. 8A  illustrates a representative received pulse signal at the input to the correlator. 
         FIG. 8B  illustrates a sequence of representative impulse signals in the correlation process. 
         FIG. 8C  illustrates the output of the correlator for each of the time offsets of FIG.  8 B. 
       FIG.  9 (A) through (D) illustrate in plan view of a variety of representative spherical monopole antennas. 
         FIG. 10  is an illustration of another embodiment of a three-dimensional monopole antenna. 
       FIG.  11 (A) through (E) illustrate representative embodiments of spheroidal dipole antenna structures. 
         FIG. 12  illustrates the relative sizes for a biconical dipole antenna (A) vis-à-vis a spheroidal dipole (B) in terms of lowest wavelength to be handled by the antenna. 
       FIG.  13 (A) illustrates a planar Vivaldi Slot exponential notch antenna in top plan view. 
       FIG.  13 (B) is a side view of the antenna illustrated in FIG.  13 (A), taken along Section B—B in FIG.  13 (A). 
         FIG. 14  is a schematic diagram of detail of an antenna feed structure for a spheroidal monopole antenna. 
         FIG. 15  is a schematic diagram of detail of an antenna feed structure for a spheroidal dipole antenna. 
       FIG.  16 (A) is a schematic section view of a spheroidal dipole antenna included within a dielectric structure. 
       FIG.  16 (B) is a schematic section view of a spheroidal monopole antenna included within a dielectric structure. 
       FIG.  17 (A) is a top plan view of an integrated circuit employment of a planar ovoidal antenna embodied in a dipole ovoidal antenna having radiating elements arrayed on opposite sides of a substrate. 
       FIG.  17 (B) is a top plan view of an integrated circuit employment of a planar ovoidal antenna embodied in a dipole ovoidal antenna having radiating elements arrayed on one side of a substrate. 
       FIG.  18 (A) is a side view of a right angle coaxial connector feed structure with a planar antenna. 
       FIG.  18 (B) is a side view of a straight coaxial connector feed structure with a planar antenna. 
       FIG.  18 (C) is a top view of a curved feed interface arrangement for an antenna of the sort illustrated in FIG.  18 (A) or FIG.  18 (B) taken along Section  18 CD— 18 CD of FIG.  18 (A) or FIG. (B). 
       FIG.  18 (D) is a top view of a straight feed interface arrangement for an antenna of the sort illustrated in FIG.  18 (A) or FIG.  18 (B) taken along Section  18 CD— 18 CD of FIG.  18 (A) or FIG. (B). 
         FIG. 19  is a perspective view in partial section of a PCMCIA card with an integral ultra wideband antenna. 
         FIG. 20  is a table summarizing performance of various antennas vis-à-vis criteria considered important for a commercially successful impulse radio communication system antenna. 
         FIG. 21  is a plan view of a schematic representation of a quadropole planar antenna according to the present invention. 
     
    
    
     DETAILED DESCRIPTION OF THE EMBODIMENTS 
     Overview of the Invention 
     The present invention will now be described more fully in detail with reference to the accompanying drawings, in which the preferred embodiments of the invention are shown. This invention should not, however, be construed as limited to the embodiments set forth herein; rather, they are provided so that this disclosure will be thorough and complete and will fully convey the scope of the invention to those skilled in art. Like numbers refer to like elements throughout. 
     Impulse Radio Basics 
     This section is directed to technology basics and provides the reader with an introduction to impulse radio concepts, as well as other relevant aspects of communications theory. This section includes subsections relating to waveforms, pulse trains, coding for energy smoothing and channelization, modulation, reception and demodulation, interference resistance, processing gain, capacity, multipath and propagation, distance measurement, and qualitative and quantitative characteristics of these concepts. It should be understood that this section is provided to assist the reader with understanding the present invention, and should not be used to limit the scope of the present invention. 
     Impulse radio refers to a radio system based on short, low duty cycle pulses. An ideal impulse radio waveform is a short Gaussian monocycle. As the name suggests, this waveform attempts to approach one cycle of radio frequency (RF) energy at a desired center frequency. Due to implementation and other spectral limitations, this waveform may be altered significantly in practice for a given application. Most waveforms with enough bandwidth approximate a Gaussian shape to a useful degree. 
     Impulse radio can use many types of modulation, including AM, time shift (also referred to as pulse position) and M-ary versions. The time shift method has simplicity and power output advantages that make it desirable. In this document, the time shift method is used as an illustrative example. 
     In impulse radio communications, the pulse-to-pulse interval can be varied on a pulse-by-pulse basis by two components: an information component and a pseudo-random code component. Generally, conventional spread spectrum systems make use of pseudo-random codes to spread the normally narrow band information signal over a relatively wide band of frequencies. A conventional spread spectrum receiver correlates these signals to retrieve the original information signal. Unlike conventional spread spectrum systems, the pseudo-random code for impulse radio communications is not necessary for energy spreading because the monocycle pulses themselves have an inherently wide bandwidth. Instead, the pseudo-random code is used for channelization, energy smoothing in the frequency domain, resistance to interference, and reducing the interference potential to nearby receivers. 
     The impulse radio receiver is typically a direct conversion receiver with a cross correlator front end in which the front end coherently converts an electromagnetic pulse train of monocycle pulses to a baseband signal in a single stage. The baseband signal is the basic information signal for the impulse radio communications system. It is often found desirable to include a subcarrier with the baseband signal to help reduce the effects of amplifier drift and low frequency noise. The subcarrier that is typically implemented alternately reverses modulation according to a known pattern at a rate faster than the data rate. This same pattern is used to reverse the process and restore the original data pattern just before detection. This method permits alternating current (AC) coupling of stages, or equivalent signal processing to eliminate direct current (DC) drift and errors from the detection process. This method is described in detail in U.S. Pat. No. 5,677,927 to Fullerton et al. 
     In impulse radio communications utilizing time shift modulation, each data bit typically time position modulates many pulses of the periodic timing signal. This yields a modulated, coded timing signal that comprises a train of identically shaped pulses for each single data bit. The impulse radio receiver integrates multiple pulses to recover the transmitted information. 
     Waveforms 
     Impulse radio refers to a radio system based on short, low duty cycle pulses. In the widest bandwidth embodiment, the resulting waveform approaches one cycle per pulse at the center frequency. In more narrow band embodiments, each pulse consists of a burst of cycles usually with some spectral shaping to control the bandwidth to meet desired properties such as out of band emissions or in-band spectral flatness, or time domain peak power or burst off time attenuation. 
     For system analysis purposes, it is convenient to model the desired waveform in an ideal sense to provide insight into the optimum behavior for detail design guidance. One such waveform model that has been useful is the Gaussian monocycle as shown in FIG.  1 A. This waveform is representative of the transmitted pulse produced by a step function into an ultra wideband antenna. The basic equation normalized to a peak value of 1 is as follows: 
           f   mono     ⁡     (   t   )       =       e     ⁢     (     t   σ     )     ⁢     e       -     t   2         2   ⁢     σ   2                 
 
Where,
         σ is a time scaling parameter,   t is time,   f mono (t) is the waveform voltage, and   e is the natural logarithm base.       

     The frequency domain spectrum of the above waveform is shown in FIG.  1 B. The corresponding equation is:
 
 F   mono ( f )=(2π) 3/2   σfe   −2(πσf)     2   
 
     The center frequency (f c ), or frequency of peak spectral density is: 
         f   c     =     1     2   ⁢   πσ           
 
     These pulses, or bursts of cycles, may be produced by methods described in the patents referenced above or by other methods that are known to one of ordinary skill in the art. Any practical implementation will deviate from the ideal mathematical model by some amount. In fact, this deviation from ideal may be substantial and yet yield a system with acceptable performance. This is especially true for microwave implementations, where precise waveform shaping is difficult to achieve. These mathematical models are provided as an aid to describing ideal operation and are not intended to limit the invention. In fact, any burst of cycles that adequately fills a given bandwidth and has an adequate on-off attenuation ratio for a given application will serve the purpose of this invention. 
     A Pulse Train 
     Impulse radio systems can deliver one or more data bits per pulse; however, impulse radio systems more typically use pulse trains, not single pulses, for each data bit. As described in detail in the following example system, the impulse radio transmitter produces and outputs a train of pulses for each bit of information. 
     Prototypes have been built having pulse repetition frequencies including 0.7 and 10 megapulses per second (Mpps, where each megapulse is 10 6  pulses).  FIGS. 2A and 2B  are illustrations of the output of a typical 10 Mpps system with uncoded, unmodulated, 0.5 nanosecond (ns) pulses  102 .  FIG. 2A  shows. a time domain representation of this sequence of pulses  102 .  FIG. 2B , which shows 60 MHz at the center of the spectrum for the waveform of  FIG. 2A , illustrates that the result of the pulse train in the frequency domain is to produce a spectrum comprising a set of lines  204  spaced at the frequency of the 10 Mpps pulse repetition rate. When the full spectrum is shown, the envelope of the line spectrum follows the curve of the single pulse spectrum  104  of FIG.  1 B. For this simple uncoded case, the power of the pulse train is spread among roughly two hundred comb lines. Each comb line thus has a small fraction of the total power and presents much less of an interference problem to receiver sharing the band. 
     It can also be observed from  FIG. 2A  that impulse radio systems typically have very low average duty cycles resulting in average power significantly lower than peak power. The duty cycle of the signal in the present example is 0.5%, based on a 0.5 ns pulse in a 100 ns interval. 
     Coding for Energy Smoothing and Channelization 
     For high pulse rate systems, it may be necessary to more finely spread the spectrum than is achieved by producing comb lines. This may be done by pseudo-randomly positioning each pulse relative to its nominal position. 
       FIG. 3  is a plot illustrating the impact of a pseudo-noise (PN) code dither on energy distribution in the frequency domain (A pseudo-noise, or PN code is a set of time positions defining the pseudo-random positioning for each pulse in a sequence of pulses).  FIG. 3 , when compared to  FIG. 2B , shows that the impact of using a PN code is to destroy the comb line structure and spread the energy more uniformly. This structure typically has slight variations which are characteristic of the specific code used. 
     The PN code also provides a method of establishing independent communication channels using impulse radio. PN codes can be designed to have low cross correlation such that a pulse train using one code will seldom collide on more than one or two pulse positions with a pulses train using another code during any one data bit time. Since a data bit may comprise hundreds of pulses, this represents a substantial attenuation of the unwanted channel. 
     Modulation 
     Any aspect of the waveform can be modulated to convey information. Amplitude modulation, phase modulation, frequency modulation, time shift modulation and M-ary versions of these have been proposed. Both analog and digital forms have been implemented. Of these, digital time shift modulation has been demonstrated to have various advantages and can be easily implemented using a correlation receiver architecture. 
     Digital time shift modulation can be implemented by shifting the coded time position by an additional amount (that is, in addition to PN code dither) in response to the information signal. This amount is typically very small relative to the PN code shift. In a 10 Mpps system with a center frequency of 2 GHz., for example, the PN code may command pulse position variations over a range of 100 ns; whereas, the information modulation may only deviate the pulse position by 150 ps. 
     Thus, in a pulse train of n pulses, each pulse is delayed a different amount from its respective time base clock position by an individual code delay amount plus a modulation amount, where n is the number of pulses associated with a given data symbol digital bit. 
     Modulation further smooths the spectrum, minimizing structure in the resulting spectrum. 
     Reception and Demodulation 
     Clearly, if there were a large number of impulse radio users within a confined area, there might be mutual interference. Further, while the PN coding minimizes that interference, as the number of users rises, the probability of an individual pulse from one user&#39;s sequence being received simultaneously with a pulse from another user&#39;s sequence increases. Impulse radios are able to perform in these environments, in part, because they do not depend on receiving every pulse. The impulse radio receiver performs a correlating, synchronous receiving function (at the RF level) that uses a statistical sampling and combining of many pulses to recover the transmitted information. 
     Impulse radio receivers typically integrate from 1 to 1000 or more pulses to yield the demodulated output. The optimal number of pulses over which the receiver integrates is dependent on a number of variables, including pulse rate, bit rate, interference levels, and range. 
     Interference Resistance 
     Besides channelization and energy smoothing, the PN coding also makes impulse radios highly resistant to interference from all radio communications systems, including other impulse radio transmitters. This is critical as any other signals within the band occupied by an impulse signal potentially interfere with the impulse radio. Since there are currently no unallocated bands available for impulse systems, they must share spectrum with other conventional radio systems without being adversely affected. The PN code helps impulse systems discriminate between the intended impulse transmission and interfering transmissions from others. 
       FIG. 4  illustrates the result of a narrow band sinusoidal interference signal  402  overlaying an impulse radio signal  404 . At the impulse radio receiver, the input to the cross correlation would include the narrow band signal  402 , as well as the received ultra wideband impulse radio signal  404 . The input is sampled by the cross correlator with a PN dithered template signal  406 . Without PN coding, the cross correlation would sample the interfering signal  402  with such regularity that the interfering signals could cause significant interference to the impulse radio receiver. However, when the transmitted impulse signal is encoded with the PN code dither (and the impulse radio receiver template signal  406  is synchronized with that identical PN code dither) the correlation samples the interfering signals pseudo-randomly. The samples from the interfering signal add incoherently, increasing roughly according to square root of the number of samples integrated; whereas, the impulse radio samples add coherently, increasing directly according to the number of samples integrated. Thus, integrating over many pulses overcomes the impact of interference. 
     Processing Gain 
     Impulse radio is resistant to interference because of its large processing gain. For typical spread spectrum systems, the definition of processing gain, which quantifies the decrease in channel interference when wide-band communications are used, is the ratio of the bandwidth of the channel to the bit rate of the information signal. For example, a direct sequence spread spectrum system with a 10 kHz information bandwidth and a 10 MHz channel bandwidth yields a processing gain of 1000 or 30 dB. However, far greater processing gains are achieved with impulse radio systems, where for the same 10 kHz information bandwidth is spread across a much greater 2 GHz. channel bandwidth, the theoretical processing gain is 200,000 or 53 dB. 
     Capacity 
     It has been shown theoretically, using signal to noise arguments, that thousands of simultaneous voice channels are available to an impulse radio system as a result of the exceptional processing gain, which is due to the exceptionally wide spreading bandwidth. 
     For a simplistic user distribution, with N interfering users of equal power equidistant from the receiver, the total interference signal to noise ratio as a result of these other users can be described by the following equation: 
         V   tot   2     =       N   ⁢           ⁢     σ   2         Z           
 
Where
         V 2   tot  is the total interference signal to noise ratio variance, at the receiver;   N is the number of interfering users;   σ 2  is the signal to noise ratio variance resulting from one of the interfering signals with a single pulse cross correlation; and   Z is the number of pulses over which the receiver integrates to recover the modulation.       

     This relationship suggests that link quality degrades gradually as the number of simultaneous users increases. It also shows the advantage of integration gain. The number of users that can be supported at the same interference level increases by the square root of the number of pulses integrated. 
     Multipath and Propagation 
     One of the striking advantages of impulse radio is its resistance to multipath fading effects. Conventional narrow band systems are subject to multipath through the Rayleigh fading process, where the signals from many delayed reflections combine at the receiver antenna according to their seemingly random relative phases. This results in possible summation or possible cancellation, depending on the specific propagation to a given location. This situation occurs where the direct path signal is weak relative to the multipath signals, which represents a major portion of the potential coverage of a radio system. In mobile systems, this results in wild signal strength fluctuations as a function of distance traveled, where the changing mix of multipath signals results in signal strength fluctuations for every few feet of travel. 
     Impulse radios, however, can be substantially resistant to these effects. Impulses arriving from delayed multipath reflections typically arrive outside of the correlation time and thus can be ignored. This process is described in detail with reference to  FIGS. 5A and 5B . In  FIG. 5A , three propagation paths are shown. The direct path representing the straight line distance between the transmitter and receiver is the shortest. Path  1  represents a grazing multipath reflection, which is very close to the direct path. Path  2  represents a distant multipath reflection. Also shown are elliptical (or, in space, ellipsoidal) traces that represent other possible locations for reflections with the same time delay. 
       FIG. 5B  represents a time domain plot of the received waveform from this multipath propagation configuration. This figure comprises three doublet pulses as shown in FIG.  1 A. The direct path signal is the reference signal and represents the shortest propagation time. The path  1  signal is delayed slightly and actually overlaps and enhances the signal strength at this delay value. Note that the reflected waves are reversed in polarity. The path  2  signal is delayed sufficiently that the waveform is completely separated from the direct path signal. If the correlator template signal is positioned at the direct path signal, the path  2  signal will produce no response. It can be seen that only the multipath signals resulting from very close reflectors have any effect on the reception of the direct path signal. The multipath signals delayed less than one quarter wave (one quarter wave is about 1.5 inches, or 3.5cm at 2 GHz center frequency) are the only multipath signals that can attenuate the direct path signal. This region is equivalent to the first Fresnel zone familiar to narrow band systems designers. Impulse radio, however, has no further nulls in the higher Fresnel zones. The ability to avoid the highly variable attenuation from multipath gives impulse radio significant performance advantages. 
       FIG. 5A  illustrates a typical multipath situation, such as in a building, where there are many reflectors  5 A 04 ,  5 A 05  and multiple propagation paths  5 A 02 ,  5 A 01 . In this figure, a transmitter TX  5 A 06  transmits a signal which propagates along the multiple propagation paths  5 A 02 ,  5 A 04  to receiver RX  5 A 08 , where the multiple reflected signals are combined at the antenna. 
       FIG. 5B  illustrates a resulting typical received composite pulse waveform resulting from the multiple reflections and multiple propagation paths  5 A 01 ,  5 A 02 . In this figure, the direct path signal  5 A 01  is shown as the first pulse signal received. The multiple reflected signals (“multipath signals”, or “multipath”) comprise the remaining response as illustrated. 
       FIGS. 5C ,  5 D, and  5 E represent the received signal from a TM-UWB transmitter in three different multipath environments. These figures are not actual signal plots, but are hand drawn plots approximating typical signal plots.  FIG. 5C  illustrates the received signal in a very low multipath environment. This may occur in a building where the receiver antenna is in the middle of a room and is one meter from the transmitter. This may also represent signals received from some distance, such as 100 meters, in an open field where there are no objects to produce reflections. In this situation, the predominant pulse is the first received pulse and the multipath reflections are too weak to be significant.  FIG. 5D  illustrates an intermediate multipath environment. This approximates the response from one room to the next in a building. The amplitude of the direct path signal is less than in FIG.  5 C and several reflected signals are of significant amplitude. (Note that the scale has been increased to normalize the plot.)  FIG. 5E  approximates the response in a severe multipath environment such as: propagation through many rooms; from corner to corner in a building; within a metal cargo hold of a ship; within a metal truck trailer; or within an intermodal shipping container. In this scenario, the main path signal is weaker than in FIG.  5 D. (Note that the scale has been increased again to normalize the plot.) In this situation, the direct path signal power is small relative to the total signal power from the reflections. 
     An impulse radio receiver in accordance with the present invention can receive the signal and demodulate the information using either the direct path signal or any multipath signal peak having sufficient signal to noise ratio. Thus, the impulse radio receiver can select the strongest response from among the many arriving signals. In order for the signals to cancel and produce a null at a given location, dozens of reflections would have to be cancelled simultaneously and precisely while blocking the direct path—a highly unlikely scenario. This time separation of multipath signals together with time resolution and selection by the receiver permit a type of time diversity that virtually eliminates cancellation of the signal. In a multiple correlator rake receiver, performance is further improved by collecting the signal power from multiple signal peaks for additional signal to noise performance. 
     Where the system of  FIG. 5A  is a narrow band system and the delays are small relative to the data bit time, the received signal is a sum of a large number of sine waves of random amplitude and phase. In the idealized limit, the resulting envelope amplitude has been shown to follow a Rayleigh probability distribution as follows: 
         p   ⁡     (   r   )       =       1     σ   2       ⁢     exp   ⁡     (       -     r   2         2   ⁢     σ   2         )             
 
where
         r is the envelope amplitude of the combined multipath signals, and   2σ 2  is the RMS power of the combined multipath signals.       

     This distribution shown in FIG.  5 F. It can be seen in  FIG. 5F  that 10% of the time, the signal is more than 16 dB attenuated. This suggests that 16 dB fade margin is needed to provide 90% link availability. Values of fade margin from 10 to 40 dB have been suggested for various narrow band systems, depending on the required reliability. This characteristic has been the subject of much research and can be partially improved by such techniques as antenna and frequency diversity, but these techniques result in additional complexity and cost. 
     In a high multipath environment such as inside homes, offices, warehouses, automobiles, trailers, shipping containers, or outside in the urban canyon or other situations where the propagation is such that the received signal is primarily scattered energy, impulse radio, according to the present invention, can avoid the Rayleigh fading mechanism that limits performance of narrow band systems. This is illustrated in  FIG. 5G and 5H  in a transmit and receive system in a high multipath environment  5 G 00 , wherein the transmitter  5006  transmits to receiver  5 G 08  with the signals reflecting off reflectors  5 G 03  which form multipaths  5 G 02 . The direct path is illustrated as  5 G 01  with the signal graphically illustrated at  5 H 02  with the vertical axis being the signal strength in volts and horizontal axis representing time in nanoseconds. Multipath signals are graphically illustrated at  5 H 04 . 
     Distance Measurement and Position Location 
     Impulse systems can measure distances to extremely fine resolution because of the absence of ambiguous cycles in the waveform. Narrow band systems, on the other hand, are limited to the modulation envelope and cannot easily distinguish precisely which RF cycle is associated with each data bit because the cycle-to-cycle amplitude differences are so small they are masked by link or system noise. Since the impulse radio waveform has no multi-cycle ambiguity, this allows positive determination of the waveform position to less than a wavelength—potentially, down to the noise floor of the system. This time position measurement can be used to measure propagation delay to determine link distance, and once link distance is known, to transfer a time reference to an equivalently high degree of precision. Systems have been built that have shown the potential for centimeter distance resolution, which is equivalent to about 30 ps of time transfer resolution. See, for example, commonly owned, co-pending U.S. patent applications Ser. No. 09/045,929, filed Mar. 23, 1998, titled “Ultrawide-Band Position Determination System and Method”, and U.S. patent application Ser. No. 09/083,993, filed May 26, 1998, titled “System and Method for Distance Measurement by Inphase and Quadrature Signals in a Radio System”, both of which are incorporated herein by reference. Finally, distance measuring and position location using impulse radio using a plurality of distance architectures is enabled in co-pending and commonly owned U.S. patent application Ser. No. 09/456,409, filed Dec. 8, 1999, titled, “System and Method for Person or Object Position Location Utilizing Impulse Radio.” 
     Exemplary Transceiver Implementation Transmitter 
     An exemplary embodiment of an impulse radio transmitter  602  of an impulse radio communication system having one subcarrier channel will now be described with reference to FIG.  6 . 
     The transmitter  602  comprises a time base  604  that generates a periodic timing signal  606 . The time base  604  typically comprises a voltage controlled oscillator (VCO), or the like, having a high timing accuracy and low jitter, on the order of picoseconds (Ps). The voltage control to adjust the VCO center frequency is set at calibration to the desired center frequency used to define the transmitter&#39;s nominal pulse repetition rate. The periodic timing signal  606  is supplied to a precision timing generator  608 . 
     The precision timing generator  608  supplies synchronizing signals  610  to the code source  612  and utilizes the code source output  614  together with an internally generated subcarrier signal (which is optional) and an information signal  616  to generate a modulated, coded timing signal  618 . 
     The code source  612  comprises a storage device such as a random access memory (RAM), read only memory (ROM), or the like, for storing suitable PN codes and for outputting the PN codes as a code signal  614 . Alternatively, maximum length shift registers or other computational means can be used to generate the PN codes. 
     An information source  620  supplies the information signal  616  to the precision timing generator  608 . The information signal  616  can be any type of intelligence, including digital bits representing voice, data, imagery, or the like, analog signals, or complex signals. 
     A pulse generator  622  uses the modulated, coded timing signal  618  as a trigger to generate output pulses. The output pulses are sent to a transmit antenna  624  via a transmission line  626  coupled thereto. The output pulses are converted into propagating electromagnetic pulses by the transmit antenna  624 . In the present embodiment, the electromagnetic pulses are called the emitted signal, and propagate to an impulse radio receiver  702 , such as shown in  FIG. 7 , through a propagation medium, such as air, in a radio frequency embodiment. In a preferred embodiment, the emitted signal is wide-band or ultra wideband, approaching a monocycle pulse as in FIG.  1 A. However, the emitted signal can be spectrally modified by filtering of the pulses. This filtering will usually cause each monocycle pulse to have more zero crossings (more cycles) in the time domain. In this case, the impulse radio receiver can use a similar waveform as the template signal in the cross correlator for efficient conversion. 
     Receiver 
     An exemplary embodiment of an impulse radio receiver  702  (hereinafter called the receiver) for the impulse radio communication system is now described with reference to FIG.  7 . More specifically, the system illustrated in  FIG. 7  is for reception of digital data wherein one or more pulses are transmitted for each data bit. 
     The receiver  702  comprises a receive antenna  704  for receiving a propagated impulse radio signal  706 . A received signal  708  from the receive antenna  704  is coupled to a cross correlator or sampler  710  to produce a baseband output  712 . The cross correlator or sampler  710  includes multiply and integrate functions together with any necessary filters to optimize signal to noise ratio. 
     The receiver  702  also includes a precision timing generator  714 , which receives a periodic timing signal  716  from a receiver time base  718 . This time base  718  is adjustable and controllable in time, frequency, or phase, as required by the lock loop in order to lock on the received signal  708 . The precision timing generator  714  provides synchronizing signals  720  to the code source  722  and receives a code control signal  724  from the code source  722 . The precision timing generator  714  utilizes the periodic timing signal  716  and code control signal  724  to produce a coded timing signal  726 . The template generator  728  is triggered by this coded timing signal  726  and produces a train of template signal pulses  730  ideally having waveforms substantially equivalent to each pulse of the received signal  708 . The code for receiving a given signal is the same code utilized by the originating transmitter  602  to generate the propagated signal  706 . Thus, the timing of the template pulse train  730  matches the timing of the received signal pulse train  708 , allowing the received signal  708  to be synchronously sampled in the correlator  710 . The correlator  710  ideally comprises a multiplier followed by a short term integrator to sum the multiplier product over the pulse interval. Further examples and details of correlation and sampling processes can be found in commonly owned U.S. Pat. Nos. 4,641,317, 4,743,906, 4,813,057 and 4,979,186 which are incorporated herein by reference, and commonly owned and co-pending application Ser. No. 09/356,384, filed Jul. 16, 1999, titled: “Baseband Signal Converter Device for a Wideband Impulse Radio Receiver,” which is incorporated herein by reference. 
     The output of the correlator  710 , also called a baseband signal  712 , is coupled to a subcarrier demodulator  732 , which demodulates the subcarrier information signal from the subcarrier. The purpose of the optional subcarrier process, when used, is to move the information signal away from DC (zero frequency) to improve immunity to low frequency noise and offsets. The output of the subcarrier demodulator  732  is then filtered or integrated in a pulse summation stage  734 . The pulse summation stage produces an output representative of the sum of a number of pulse signals comprising a single data bit. The output of the pulse summation stage  734  is then compared with a nominal zero (or reference) signal output in a detector stage  738  to determine an output signal  739  representing an estimate of the original information signal  616 . 
     The baseband signal  712  is also input to a lowpass filter  742  (also referred to as lock loop filter  742 ). A control loop comprising the lowpass filter  742 , time base  718 , precision timing generator  714 , template generator  728 , and correlator  710  is used to generate a filtered error signal  744 . The filtered error signal  744  provides adjustments to the adjustable time base  718  to time position the periodic timing signal  726  in relation to the position of the received signal  708 . 
     In a transceiver embodiment, substantial economy can be achieved by sharing part or all of several of the functions of the transmitter  602  and receiver  702 . Some of these include the time base  718 , precision timing generator  714 , code source  722 , antenna  704 , and the like. 
       FIGS. 8A-8C  illustrate the cross correlation process and the correlation function.  FIG. 8A  shows the waveform of a template signal.  FIG. 8B  shows the waveform of a received impulse radio signal at a set of several possible time offsets.  FIG. 8C  represents the output of the correlator (multiplier and short time integrator) for each of the time offsets of FIG.  8 B. Thus, this graph,  FIG. 8C , does not show a waveform that is a function of time, but rather a function of time-offset, i.e., for any given pulse received, there is only one corresponding point which is applicable on this graph. This is the point corresponding to the time offset of the template signal used to receive that pulse. 
     Further examples and details of subcarrier processes and precision timing can be found described in U.S. Pat. No. 5,677,927, titled “An Ultrawide-Band Communications System and Method”, and commonly owned co-pending application Ser. No. 09/146,524, filed Sep. 3, 1998, titled “Precision Timing Generator System and Method”, both of which are incorporated herein by reference. 
     Impulse Radio as Used in the Present Invention 
     When utilized in a radio communication network, the characteristics of impulse radio significantly improve the state of the art. The present invention is particularly valuable when used in a radio network employing impulse radio; the present invention is compact and exhibits efficient omni-directional non-dispersive radio transmission and receive characteristics with reduced ringing in the presence of impulse signals. 
     Detailed Description of the Preferred Embodiment 
     A vital component for any radio communication system is the antenna system or systems employed for transmitting and receiving radio frequency (RF) signals. Generally, characteristics that relate to good transmitting quality for a particular antenna apply with equal relevance to receiving characteristics of the antenna. Characteristics that are preferably optimized for antennas employed with an impulse radio communication system are that the antennas should be a broadband antenna that is small and compact, well-matched (preferably impedance-matched with a 50 ohm load), efficient without a propensity for ringing when subjected to pulsed signals, non-dispersive in its transceiving operations, and omni-directional. From a practical standpoint, an antenna system should be easy to make with reliable quality in production volumes (as contrasted with volumes appropriate for prototype manufacture). 
     Spheroidal antennas—dipoles, and monopoles—have been found by the inventor to be well matched, efficient, physically small, and radiate non-dispersively and omni-directionally. A significant shortcoming of such antennas, however, is that they are relatively expensive and difficult to manufacture, especially in production level numbers. 
     A monopole spheroidal antenna consists of a single spheroidal radiating element, mounted on a ground plane combined with a feed structure. Any of the spheroidal dipole antennas has a monopole analog obtained by driving the upper radiating element against a ground plane. (Radiating elements may be referred to as “upper” and “lower,” but this is only a naming convention and should not be considered as limiting in any way the orientation of the antenna.). Although a solid spheroid or spheroidal shell may be preferred, excellent results may also be obtained by a mesh, a sparse wire configuration, or a collection of plates embodiment of the antenna. 
     FIG.  9 (A) through (D) illustrate in plan view of a variety of representative spherical monopole antennas. In FIG.  9 (A), a monopole antenna  900  includes a feed structure  902  coupled with a radiating (or receiving) structure  904  and a ground plane  906 . Radiating structure  904  may be in the shape of a sphere, or in a spheroidal shape. Further, radiating structure  904  may be a solid sphere or spheroid, or the desired sphere or spheroid shape may be provided by a mesh, a sparse wire configuration or a collection of plates, so long as the outer perimeter circumscribed by the radiating element or elements is the desired sphere or spheroid shape. The interior of radiating structure  904  may contain a microphone, earphone or another desired electrical, mechanical or electro-mechanical device so long as an interface through radiating structure  904  is placed appropriately to minimize interference with transmission or reception by radiating structure  904 . 
     In FIG.  9 (A), ground plane  906  is a substantially circular shape with radiating element  904  situated substantially centrally within ground plane  906 . In some applications using particular antenna systems one may wish to alter the pattern of radiation (or the pattern of reception) or reduce signal diffraction. One structural approach to achieving such results is to treat the edge of the ground plane associated with the monopole antenna. Alternate representative embodiments are illustrated in FIGS.  9 (B),  9 (C), and  9 (D). Thus, in FIG.  9 (B), a monopole antenna  900 B includes a feed structure  902 B connected with a radiating element  904 B and a ground plane  906 B. Ground plane  906 B is a substantially circular planar element with radiating element  904 B substantially centrally located therein. Ground plane  906 B includes a rolled edge  908 B rolled away from radiating element  904 B. In FIG.  9 (C), a monopole antenna  900 C includes a feed structure  902 C connected with a radiating element  904 C and a ground plane  906 C. Ground plane  906 C is a substantially circular shape with radiating element  904 C situated substantially centrally within ground plane  906 C. Ground plane  906 C includes an interrupted, serrated and rolled edge  908 C resulting in a “daisy” patterned rolled edge turning away from radiating element  904 C. In FIG.  9 (D), a monopole antenna  900 D includes a feed structure  902 D connected with a radiating element  904 D and a ground plane  906 D. A conical ground plane may be used to expand the field of view of the radiating element. Ground plane  906 D is a substantially conical shape with radiating element  904 D situated substantially at the apex  903 D of ground plane  906 D. 
     The various spheroidal monopole antenna structures illustrate representative monopole antenna structures that may be employed in an impulse radio system. Combinations of the structures illustrated in FIG.  9 (A)-(D), or combinations employing different structures may be selected by an antenna system designer for particular antenna characteristics, such as varying diffraction patterns, varying radiating patterns or other reasons known to a skilled antenna designer. Preferably, ground plane  900 ,  900 A,  900 B,  900 C,  900 D should have a radius at least approximately equal to the height of the spheroid, or larger. By way of further example, a monopole radiating element  904 ,  904 B,  904 C,  904 D may be mounted to an area on the exterior of a vehicle or to an equipment chassis or housing if the area forms a suitable conducting ground plane of sufficient size. 
       FIG. 10  is an illustration of another embodiment of a three-dimensional monopole antenna. In  FIG. 10 , a “volcano smoke” antenna  1000 , as described by Kraus (John D. Kraus,  Antennas,  2 nd  edition; New York: McGraw Hill, 1988) includes a tapered feed structure  1012  joined with a spheroidal radiating element  1014  in a smooth transition region  1010 . A backplane  1016  is smoothly undulating to effect radiating (and receiving) patterns as desired. “Volcano smoke” antenna  1000  may exhibit acceptable broadband characteristics for impulse radio applications; use of such antennas in impulse radio applications is not known to have occurred. Tapered feed structure  1012  provides good matching, and radiating element  1014  radiates omni-directionally. However, transition region  1010  requires a bulky configuration for “volcano smoke”antenna  1000  that resists fabrication in a compact configuration. Further, the smooth transition required of transition region  1010  makes manufacture of “volcano smoke” antenna  1000  difficult. 
     A dipole spheroidal antenna consists of two spheroidal radiating elements arranged along an axis. Typically, but not necessarily, the feed structure is arranged along the axis of the spheroidal dipole. The feed structure connects a radio frequency (RF) transmission line to the gap between the spheroidal radiating elements. The RF transmission line is usually a coaxial transmission line or connector, but may be any of a variety of different feed structures, including stripline or twin lead structures. Advantages similar to the advantages enjoyed by a solid structure may be obtained in large part using a mesh, wire frame or a plate or fin type element that occupies a spheroidal volume. Even a relatively sparse mesh, wire frame or grid provides substantially full measure of the benefits realized using a solid structure. In practicality, such a mesh, wire frame or grid structure may be preferred for advantages offered over a solid structure in terms of manufacturability, economies of material and weight, lesser wind loading, or improved aesthetic considerations. 
     FIG.  11 (A) through (E) illustrate representative embodiments of spheroidal dipole antenna structures. The illustrations of  FIG.11  (A) through (E) are intended to indicate the respective antenna structures as solid configurations, as wire mesh configurations circumscribing a desired shape or to indicate the desired shape that is circumscribed by a multiple fin or grid structure. 
     In FIG.  11 (A), a spherical dipole antenna  1100 A includes a feed structure  1102 A, a first spherical radiating (or receiving) element  1104 A and a second spherical radiating (or receiving) element  1106 A. Spherical radiating elements  1104 A,  1106 A are oriented substantially symmetrically about an axis  1108 A. Axis  1108 A is typically coaxial with feed structure  1102 A, but such an axial coincidence is not required. Feed structure  1102 A is connected (not visible in FIG.  11 (A)) with spherical radiating elements  1104 A,  1106 A appropriately to establish a radio frequency antenna structure. 
     In FIG.  11 (B), a prolate spheroidal dipole antenna  1100 B includes a feed structure  1102 B, a first spheroidal radiating (or receiving) element  1104 B and a second spheroidal radiating (or receiving) element  1106 B. Spheroidal radiating elements  1104 B,  1106 B each have a major axis and a minor axis and are each oriented with their respective major axis substantially coincident with an axis  1108 B. Axis  1108 B is typically coaxial with feed structure  1102 B, but such an axial coincidence is not required. Feed structure  1102 B is connected (not visible in FIG.  11 (B)) with spheroidal radiating elements  1104 B,  1106 B appropriately to establish a radio frequency antenna structure. 
     In FIG.  11 (C), an oblate spheroidal dipole antenna  1100 C includes a feed structure  1102 C, a first spheroidal radiating (or receiving) element  1104 C and a second spheroidal radiating (or receiving) element  1106 C. Spheroidal radiating elements  1104 C,  1106 C each have a major axis and a minor axis and are each oriented with their respective minor axis substantially coincident with an axis  1108 C. Axis  1108 C is typically coaxial with feed structure  1102 C, but such an axial coincidence is not required. Feed structure  1102 C is connected (not visible in FIG.  11 (C)) with spheroidal radiating elements  1104 C,  1106 C appropriately to establish a radio frequency antenna structure. 
     In FIG.  11 (D), a Blefuscuan spheroidal dipole antenna  1100 D includes a feed structure  1102 D, a first spheroidal radiating (or receiving) element  1104 D and a second spheroidal radiating (or receiving) element  1106 D. (An ovoid excited from the smaller end is referred to as a “Lilliputian ovoid” after the miniature people described in Gulliver&#39;s Travels who ate their eggs from the small end. An ovoid excited from the fatter end is referred to as a “Blefuscuan ovoid” after the hereditary rivals of the Lilliputians who ate their eggs from the fat end.) Radiating elements  1104 D,  1106 D are each egg-shaped with a major axis, and are each oriented with their respective major axis substantially coincident with an axis  1108 D. Axis  1108 D is typically coaxial with feed structure  1102 D, but such an axial coincidence is not required. Feed structure  1102 D is connected (not visible in FIG.  11 (D)) with spheroidal radiating elements  1104 D,  1106 D appropriately to establish a radio frequency antenna structure. 
     In FIG.  11 (E), a Lilliputian spheroidal dipole antenna  1100 E includes a feed structure  1102 E, a first spheroidal radiating (or receiving) element  1104 E and a second spheroidal radiating (or receiving) element  1106 E. Radiating elements  1104 E,  1106 E are each egg-shaped with a major axis, and are each oriented with their respective major axis substantially coincident with an axis  1108 E. Axis  1108 E is typically coaxial with feed structure  1102 E, but such an axial coincidence is not required. Feed structure  1102 E is connected (not visible in FIG.  11 (E)) with spheroidal radiating elements  1104 E,  1106 E appropriately to establish a radio frequency antenna structure. 
       FIG. 12  illustrates the relative sizes for a biconical dipole antenna (A) vis-à-vis a spheroidal dipole (B) in terms of lowest wavelength to be handled by the antenna. In FIG.  12 (A), a biconical dipole antenna  1200 A with an included angle of 120° will have a 100Ω input with a voltage standing wave ratio (VSWR) of &lt;2:1, over a 6:1 bandwidth. The geometry of biconical dipole antenna  1200 A requires a minimum diameter equal to the wavelength of the lower frequency limit λ L  of biconical dipole antenna  1200 A. Accordingly, biconical dipole antenna  1200 A would have a height of 0.577λ L . Antennas as large as this are prone to dispersive operation; that is, they radiate (or receive) different frequencies from different regions of the antenna. Such a large antenna would not be amenable for use with a small hand held portable device. 
     At the lower frequency wavelength limit λ L  of spheroidal dipole antenna  1200 B, diameter of radiating elements needs only to be in the range of approximately 
         λ   6     ⁢           ⁢   to   ⁢           ⁢       λ   10     .         
 
This is noteworthy because it is usually expected that radiating elements have a minimum size on the order of 
       λ   4       
 
for efficient radiation (and reception). This small diametral requirement allows spheroidal dipole antenna  1200 B to have a height of approximately 0.33λ L  Such small size helps spheroidal dipole antenna  1200 B radiate non-dispersively; such non-dispersive operation is well suited for impulse radio transmissions without ringing.
 
     In contrast to biconical dipole antenna  1200 A, spheroidal dipole antenna  1200 B can easily achieve a sufficiently favorable VSWR to yield an efficiency of 90-96% over a large bandwidth. Such excellent efficiency over a large bandwidth indicates that reflections (and, hence, ringing) are reduced in spheroidal dipole antenna  1200 B, even without resistive loading. Moreover, spheroidal dipole antenna  1200 B is easily matched to 50Ω. 
     FIG.  13 (A) illustrates a planar Vivaldi Slot exponential notch antenna in top plan view. FIG.  13 (B) is a side view of the antenna illustrated in FIG.  13 (A), taken along Section B—B in FIG.  13 (A). In FIG.  13 (A), an exponential notch antenna  1300  includes a dielectric substrate  1302  having a top side  1304  and a bottom side  1306 . A conductive material  1308  is arrayed on top side  1304  leaving an uncovered zone  1310  free of conductive material  1308 . A margin  1312  is established between uncovered zone  1310  and conductive material  1308  on top side  1304  of dielectric substrate  1302 . Margin  1312  traces an exponential curve; the particular shape of the exponential curve traced by margin  1312  determines operational characteristics of antenna  1300 . A strip of conductive material  1316  is arrayed upon bottom side  1306  of dielectric substrate  1302 . The placement and dimensions of strip of conductive material  1316  effects impedance matching of antenna  1300  with an associated feed structure (not shown in FIG.  13 ). Exponential notch antenna  1300  is an example of a planar antenna used in broadband radio applications. Such exponential notch antennas as antenna  1300  can be well matched to 50Ω, a desirable condition for radio applications, but such antennas are on the order of a wavelength long (i.e., the lower limit wavelength λ L ), as indicated by dimension λ L  in FIG.  13 (A). Antennas such as exponential notch antenna  1300  are easily and dependably manufactured, but they are very large and are directive in their operation; they are not omni-directional. 
     A spheroidal dipole is a special case of what has been called a “causal surface antenna”. See “The Energy Flow and Frequency Spectrum About Electric and Magnetic Dipoles” by Hans Gregory Schantz; Ph.D. Dissertation, University of Texas; August 1995. Planar ovoidal antennas have been found to have similar causal surface effects to those of spheroidal antennas. A causal surface is one through which there is no flow of electromagnetic energy. A “causal surface antenna” describes an antenna designed to have minimal stored or reactive energy. Such an antenna has a very low quality factor or “Q,” and thus a very broad bandwidth response. Because there is minimal reactive energy, a causal surface antenna presents a largely resistive match over the same broad band of frequencies. 
     Since the dimensions of a causal surface tend to be on the order of λ/2π=0.159λ, a causal surface antenna can be made significantly smaller than the typical λ/4=0.250λ dimension usually thought necessary to have an efficiently radiating antenna. This small size makes it easier to fit a causal surface antenna into a smaller volume efficient package, and also tends reduce ringing or dispersive behavior by the antenna. 
     When a static, electric Hertzian dipole antenna undergoes certain classes of exponential and damped exponential decays, there exists about the dipole a spherical “causal surface,” through which there is no flux of electromagnetic energy. This result suggests that a spherical antenna will have particularly good properties. The inventor has discovered that spheroidal and ovoidal antennas are well suited for employment with impulse radio systems for several technical and economic reasons. 
       FIG. 14  is a schematic diagram of detail of an antenna feed structure for a spheroidal monopole antenna. In  FIG. 14 , a spheroidal monopole antenna  1400  includes a spheroidal radiating element  1402 , a generally planar backplane structure  1404  and a feed structure  1406 . Feed structure  1406  is illustrated as a coaxial feed structure including a feed line  1408  substantially surrounded by a sleeve  1410 . A space  1412  between feed line  1408  and sleeve  1410  may be occupied by air or by a dielectric material. Illustrating feed structure  1406  as a coaxial feed structure is merely illustrative and is not intended to limit the variety of transmission lines or connectors that could be employed in constructing feed structure  1406 . A feed structure that is oriented substantially about the axis of an antenna is generally preferred because energy flow and surface currents are minimized at the axial locus. 
     Feed structure  1406  is coupled with antenna  1400  in a feed region  1415 . Feed line  1408  is connected with radiating element  1402  at a feed point  1414  within feed region  1415 , and sleeve  1410  is connected with back plane  1404  at a low potential connection locus  1416  within feed region  1415 . Low potential connection locus  1416  is preferably a ground connection. If additional mechanical strength or improved resistance to electrical breakdown is desired, dielectric material may be included in feed region  1415 . 
     Variation in overall spheroidal geometry of antenna  1400  may be accommodated without significantly affecting the performance of antenna  1400 . The inventor has learned that feed region  1415  is critical to provide good matching and minimal reflection while operating antenna  1400 . Prior art teaching has asserted that a region at which an antenna is connected with its feed should be point-like at a feed point and flare out from that feed point. The present invention incorporates an antenna feed region having a “blunt” or curved surface at a feed point, such as curved surface  1417  spanning a dimension “X” within feed region  1415 . The inventor has discovered that it is advantageous to provide an approximately spheroidal surface for connecting feed structure  1406 . Such a curved surface  1417  at an antenna feed point  1414  significantly lowers the impedance of the juncture between feed structure  1406  and radiating element  1402  at feed point  1414 , thereby providing an improved match to 50Ω that is not so easily attainable using prior art antenna feed arrangements (if such a preferred low impedance is attainable at all). 
     A gap width “G” between radiating element  1402  and backplane  1404  is established by the arrangement illustrated in FIG.  14 . Gap width G is a critical parameter that must be carefully arranged for providing best results using antenna  1400 . A gap width G approximately equal to diameter D of feed structure  1406  is a preferred starting dimension for beginning adjustments to optimize performance. 
     In exemplary antenna  1400 , feed structure  1406  embodies an energy guiding means radiating element  1402  cooperates with backplane structure  1404  to embody an energy channeling structure and feed region  1415  embodies a transition means. 
       FIG. 15  is a schematic diagram of detail of an antenna feed structure for a spheroidal dipole antenna. In  FIG. 15 , a spheroidal monopole antenna  1500  includes a first spheroidal radiating element  1502 , a second spheroidal radiating element  1504  and a feed structure  1506 . Feed structure  1506  is illustrated as a coaxial feed structure including a feed line  1508  substantially surrounded by a sleeve  1510 . A space  1512  between feed line  1508  and sleeve  1510  may be occupied by air or by a dielectric material. Illustrating feed structure  1506  as a coaxial feed structure is merely illustrative and is not intended to limit the variety of transmission lines or connectors that could be employed in constructing feed structure  1506 . A feed structure that is oriented substantially about the axis of an antenna is generally preferred because energy flow and surface currents are minimized at the axial locus. 
     Feed structure  1506  is coupled with antenna  1500  in a feed region  1515 . Feed line  1508  is connected with first radiating element  1502  at a feed point  1514  within feed region  1515 , and sleeve  1510  is connected with second radiating element  1504  at a low potential connection locus  1516  within feed region  1515 . Low potential connection locus  1516  is preferably a ground connection. If additional mechanical strength or improved resistance to electrical breakdown is desired, dielectric material may be included in feed region  1515 . 
     Variation in overall spheroidal geometry of radiating elements  1502 ,  1504  of antenna  1500  may be accommodated without significantly affecting the performance of antenna  1500 . The inventor has learned that feed region  1515  is critical to provide good matching and minimal reflection while operating antenna  1500 . Prior art teaching has asserted that a region at which an antenna is connected with its feed should be point-like at a feed point and flare out from that feed point. The present invention incorporates an antenna feed region having a “blunt” or curved surface at a feed point, such as curved surface  1517  spanning a dimension “X I ” about feed point  1514 , and curved surface  1519  spanning a dimension “X 2 ” about feed locus  1516  within feed region  1515 . The inventor has learned that it is advantageous to provide an approximately spheroidal surface for connecting feed structure  1506 . Such curved surfaces  1517 ,  1519  at an antenna feed point  1514  or an antenna feed locus  1516  significantly lower the impedance of the juncture between feed structure  1506  and radiating elements  1502 ,  1504  within feed region  1515 , thereby providing an improved match to 50Ω that is not so easily attainable using prior art antenna feed arrangements (if such a preferred low impedance is attainable at all). 
     A gap width “G” between radiating elements  1502 ,  1504  is established by the arrangement illustrated in FIG.  15 . Gap width G is a critical parameter that must be carefully arranged for providing best results using antenna  1500 . A gap width G approximately equal to diameter D of feed structure  1506  is a preferred starting dimension for beginning adjustments to optimize performance. 
     In exemplary antenna  1500 , feed structure  1506  embodies an energy guiding means, radiating elements  1502 ,  1504  cooperate to embody an energy channeling structure and feed region  1515  embodies a transition means. 
     To facilitate miniaturization, a spheroidal antenna may be coated or even encased in dielectric to have the effect of miniaturizing still further the dimensions of the antenna. The dielectric constant may be varied from the gap to the outer surface to improve matching. 
     FIG.  16 (A) is a schematic section view of a spheroidal dipole antenna included within a dielectric structure. In FIG.  16 (A), s spheroidal dipole antenna  1600  includes a first spheroidal radiating element  1602 , a second spheroidal radiating element  1604  and a feed structure  1606 . Feed structure  1606  is illustrated as a coaxial feed structure including a feed line  1608  substantially surrounded by a sleeve  1610 . A space  1612  between feed line  1608  and sleeve  1610  may be occupied by air or by a dielectric material. Illustrating feed structure  1606  as a coaxial feed structure is merely illustrative and is not intended to limit the variety of transmission lines or connectors that could be employed in constructing feed structure  1606 . A feed structure that is oriented substantially about the axis of an antenna is generally preferred because energy flow and surface currents are minimized at the axial locus. 
     Feed line  1608  is connected with first radiating element  1602  at a feed point  1614 , and sleeve  1610  is connected with second radiating element  1604  at a low potential connection locus  1616 . Low potential connection locus  1616  is preferably a ground connection. In order to provide additional mechanical strength or improved resistance to electrical breakdown, a dielectric wrap  1620  is installed substantially surrounding radiating elements  1602 ,  1604 . Preferably, dielectric wrap  1620  is a substantially solid spheroidal structure in its surrounding relationship with radiating elements  1602 ,  1604 . 
     FIG.  16 (B) is a schematic section view of a spheroidal monopole antenna included within a dielectric structure. In FIG.  16 (B), a spheroidal monopole antenna  1650  includes a spheroidal radiating element  1652 , a back plane  1654  and a feed structure  1656 . Feed structure  1656  is illustrated as a coaxial feed structure including a feed line  1658  substantially surrounded by a sleeve  1660 . A space  1662  between feed line  1658  and sleeve  1660  may be occupied by air or by a dielectric material. Illustrating feed structure  1656  as a coaxial feed structure is merely illustrative and is not intended to limit the variety of transmission lines or connectors that could be employed in constructing feed structure  1656 . A feed structure that is oriented substantially about the axis of an antenna is generally preferred because energy flow and surface currents are minimized at the axial locus. 
     Feed line  1658  is connected with radiating element  1652  at a feed point  1664 , and sleeve  1660  is connected with back plane  1654  at a low potential connection locus  1666 . Low potential connection locus  1666  is preferably a ground connection. In order to provide additional mechanical strength or improved resistance to electrical breakdown, a dielectric wrap  1670  is installed substantially surrounding radiating element  1652 . Preferably, dielectric wrap  1670  is a substantially solid hemispheroidal structure in its surrounding relationship with radiating element  1652 . 
     Keeping in mind the characteristics that are preferably optimized for an antenna employed with an impulse radio communication system (i.e., the antenna should be a broadband antenna that is small and compact, well-matched—preferably impedance-matched with a 50 ohm load, efficient without a propensity for ringing when subjected to pulsed signals, non-dispersive in its transceiving operations, and omni-directional) one must consider the ease of manufacture in reliable quantities provided by a planar antenna. 
     Ovoidal antennas—dipoles, and monopoles—have been found by the inventor to be well matched, efficient, physically small, and radiate non-dispersively and omni-directionally. A significant advantage of such antennas is that they are relatively inexpensive and easy to reliably manufacture in production level numbers. 
     A monopole ovoidal antenna consists of a single planar ovoidal radiating element, mounted on a ground plane combined with a feed structure. Any of the ovoidal dipole antennas has a monopole analog obtained by driving the ovoidal radiating element against a ground plane. Although a solid ovoid may be preferred, excellent results may also be obtained by a mesh or a sparse wire configuration embodiment of the antenna. 
     FIG.  17 (A) is a top plan view of an integrated circuit employment of a planar ovoidal antenna embodied in a dipole ovoidal antenna having radiating elements arrayed on opposite sides of a substrate. In FIG.  17 (A), a planar dipole ovoidal antenna  1700  includes a substantially planar substrate  1702 , a first ovoidal radiating element  1704  carried upon a top side of substrate  1702 , a second ovoidal radiating element  1706  carried upon a bottom side of substrate  1702  and support circuitry  1708  carried upon substrate  1702 . Support circuitry  1708  may be carried on either side of substrate  1702 . Support circuitry  1708  is carried upon the top side of substrate  1702  in FIG.  17 (A) for illustrative purposes. 
     A first circuit trace  1710  on the top side of substrate  1702  connects support circuitry  1708  with first radiating element  1704  at a connection locus  1713 . At high frequencies of the sort involved with radio frequency (RF) employment of antenna  1700 , and especially at the frequencies employed for ultra-wideband impulse radio applications, signals travel substantially on the surface of radiating element  1704  (i.e., at a shallow skin depth) from connection locus  1713  to a feed region  1715 . Support circuitry  1708  is connected with a second circuit trace  1712  (the connection is not visible in FIG.  17 (A)) and second circuit trace  1712  connects support circuitry  1708  with second radiating element  1706  on the bottom side of substrate  1702  in feed region  1715 . Feed region  1715  is effectively established in a region in which radiating elements  1704 ,  1706  are excited against each other. In fact, radiating elements  1704 ,  1706  may comprise a multi-layer structure to function as a microphone, earphone or another desired electrical, mechanical or electro-mechanical device so long as radiating elements  1704 ,  1706  remain substantially thin and so long as an interface through radiating elements  1704 ,  1706  is placed to minimize interference with transmission or reception by radiating elements  1704 ,  1706 . 
     Support circuitry  1708  may include, for example, radio frequency (RF) circuitry, a battery, switches, indicators, interface circuits, displays and other equipment or devices supporting or using antenna  1700 . 
     Variation in overall ovoidal geometry of radiating elements  1704 ,  1706  of antenna  1700  may be accommodated without significantly affecting the performance of antenna  1700 . The inventor has learned that feed region  1715  is critical to provide good matching and minimal reflection while operating antenna  1700 . The lessons of the present invention apply with substantially equal relevance to planar antennas as they apply to three-dimensional antennas (discussed above in connection with FIGS.  14 - 16 ). Prior art teaching has asserted that a region at which an antenna is connected with its feed should be point-like at a feed point and flare out from that feed point. The present invention incorporates an antenna feed region having a “blunt” or curved surface at a feed point, such as curved junctures  1717 ,  1719  spanning feed region  1715 . It is advantageous to provide an approximately ovoidal juncture for connecting circuit traces  1710 ,  1712 . The inventor has discovered that such curved junctures  1717 ,  1719  at an antenna feed region  1715  significantly lower the impedance of the juncture between feed structure embodied in circuit traces  1710 ,  1712 . Radiating elements  1704 ,  1706  within feed region  1715  thereby providing an improved match to 50Ω that is not so easily attainable using prior art antenna feed arrangements (if such a preferred low impedance is attainable at all). 
     A gap width “G” between radiating elements  1704 ,  1706  is established by the arrangement illustrated in FIG.  17 (A). Gap width G is a critical parameter that must be carefully arranged for providing best results using antenna  1700 . A gap width “G” approximately equal to the width W of circuit trace  1712  is a preferred starting dimension for beginning adjustments to optimize performance. 
     FIG.  17 (B) is a top plan view of an integrated circuit employment of a planar ovoidal antenna embodied in a dipole ovoidal antenna having radiating elements arrayed on one side of a substrate. In FIG.  17 (B), a planar dipole ovoidal antenna  1750  includes a substantially planar substrate  1752 , a first ovoidal radiating element  1754  carried upon a top side of substrate  1752 , a second ovoidal radiating element  1756  carried upon the top side of substrate  1752  (the same side of substrate  1752  that carries first radiating element  1754 ) and support circuitry  1758  carried upon substrate  1752 . Support circuitry  1758  may be carried on either side of substrate  1752 . Support circuitry  1758  is carried upon the top side of substrate  1752  in FIG.  17 (B) for illustrative purposes. 
     A first circuit trace  1760  on the top side of substrate  1752  connects support circuitry  1758  with first radiating element  1754  at a connection locus  1763 . At high frequencies of the sort involved with radio frequency (RF) employment of antenna  1750 , and especially at the frequencies employed for ultra-wideband impulse radio applications, signals travel substantially on the surface of radiating element  1754  (i.e., at a shallow skin depth) from connection locus  1763  to a feed region  1765 . Support circuitry  1758  is connected with a second circuit trace  1762  on the bottom side of substrate  1752  (the connection is not visible in FIG.  17 (B)) and second circuit trace  1762  connects support circuitry  1758  with second radiating element  1756  via the bottom side of substrate  1752  and a through-hole  1775  (or another via structure) in a feed region  1765 . 
     Support circuitry  1758  may include, for example, radio frequency (RF) circuitry, a battery, switches, indicators, interface circuits, displays and other equipment or devices supporting or using antenna  1750 . 
     Variation in overall ovoidal geometry of radiating elements  1754 ,  1756  of antenna  1750  may be accommodated without significantly affecting the performance of antenna  1750 . The inventor has learned that feed region  1765  is critical to provide good matching and minimal reflection while operating antenna  1750 . The lessons of the present invention apply with substantially equal relevance to planar antennas as they apply to three-dimensional antennas (discussed above in connection with FIGS.  14 - 16 ). Prior art teaching has asserted that a region at which an antenna is connected with its feed should be point-like at a feed point and flare out from that feed point. The present invention incorporates an antenna feed region having a “blunt” or curved surface at a feed point, such as curved junctures  1767 ,  1769  spanning feed region  1765 . It is advantageous to provide an approximately ovoidal juncture for connecting circuit traces  1760 ,  1762 . The inventor has discovered that such curved junctures  1767 ,  1769  at an antenna feed region  1765  significantly lower the impedance of the juncture between feed structure embodied in circuit traces  1760 ,  1762  and radiating elements  1754 ,  1756  within feed region  1765  thereby providing an improved match to 50Ω that is not so easily attainable using prior art antenna feed arrangements (if such a preferred low impedance is attainable at all). 
     A gap width “G” between radiating elements  1754 ,  1756  is established by the arrangement illustrated in FIG.  17 (B). Gap width G is a critical parameter that must be carefully arranged for providing best results using antenna  1750 . A gap width “G” approximately equal to the width W of circuit trace  1762  is a preferred starting dimension for beginning adjustments to optimize performance. 
     FIG.  18 (A) is a side view of a right angle coaxial connector feed structure with a planar antenna. In FIG.  18 (A), an antenna assembly  1800  includes a dielectric substrate  1802  carrying a first radiating element  1804  and a second radiating element  1806 . A coaxial connector  1808  provides a connection structure  1810  for a coaxial cable (not shown in FIG.  18 (A)), and a right-angle structure  1812 . Coaxial connector  1808  is affixed with dielectric substrate  1802  incorporating spacer structure  1814 . Spacer structure  1814  may, for example, include a plurality of nylon spacers, or another spacer structure appropriate to establish a gap dimension “X” from dielectric substrate  1802  appropriate for proper antenna operation by antenna assembly  1800 . Ground pins  1816  (only one ground pin  1816  is visible in FIG.  18 (A)) connect first radiating element  1804  with ground connectors  1818  (only one ground pin  1818  is visible in FIG.  18 (A)). Center pin  1820  connects second radiating element  1806  with the center connector wire of the coaxial cable (not shown in FIG.  18 (A)) attached using coaxial connector  1808 . 
     FIG.  18 (B) is a side view of a straight coaxial connector feed structure with a planar antenna. In FIG.  18 (B), an antenna assembly  1850  includes a dielectric substrate  1852  carrying a first radiating element  1854  and a second radiating element  1856 . A coaxial connector  1858  provides a connection structure  1860  for a coaxial cable (not shown in FIG.  18 (B)). Coaxial connector  1858  is affixed with dielectric substrate  1852  incorporating spacer structure  1864 . Spacer structure  1864  may, for example, include a plurality of nylon spacers, or another spacer structure appropriate to establish a gap dimension “X” from dielectric substrate  1852  appropriate for proper antenna operation by antenna assembly  1850 . Ground pins  1866  (only one ground pin  1866  is visible in FIG.  18 (B)) connect first radiating element  1854  with ground connectors  1868  (only one ground pin  1868  is visible in FIG.  18 (B)). Center pin  1870  connects second radiating element  1856  with the center connector wire of the coaxial cable (not shown in FIG.  18 (B)) attached using coaxial connector  1858 . 
     FIG.  18 (C) is a top view of a curved feed interface arrangement for an antenna of the sort illustrated in FIG.  18 (A) or FIG.  18 (B) taken along Section  18 CD— 18 CD of FIG.  18 (A) or FIG. (B). In FIG.  18 (C), radiating elements  1804 ,  1806  are carried upon dielectric substrate  1802 . In feed region  1815 , ground pins  1816  are connected with first radiating element  1804 , and center pin  1820  is connected with second radiating element  1806 . Connection may be effected using solder or other known connection techniques. A gap G is established between radiating elements  1804 ,  1806 . 
     FIG.  18 (D) is a top view of a straight feed interface arrangement for an antenna of the sort illustrated in FIG.  18 (A) or FIG.  18 (B) taken along Section  18 CD— 18 CD of FIG.  18 (A) or FIG. (B). In FIG.  18 (D), radiating elements  1854 ,  1856  are carried upon dielectric substrate  1852 . In feed region  1865 , ground pins  1866  are connected with first radiating element  1854 , and center pin  1870  is connected with second radiating element  1856 . Connection may be effected using solder or other known connection techniques. A gap G is established between radiating elements  1854 ,  1856 . 
       FIG. 19  is a perspective view in partial section of a PCMCIA card with an integral ultra wideband antenna. In  FIG. 19 , a PCMCIA (Personal Computer Memory Card International Association) apparatus  1900  includes a card-shaped device  1902  configured for insertion into an appropriate PCMCIA card receiver slot (not shown in  FIG. 19 ) to a full insertion locus  1904 . PCMCIA apparatus  1900  includes an integral antenna  1906 . An exemplary shape for antenna  1906  is illustrated in  FIG. 19  in the form of a cylindrical substrate  1908  carrying a first radiating (or receiving) element  1910  and a second radiating (or receiving) element  1912 . Radiating elements  1910 ,  1912  are fed by a feed structure  1914  (such as a transmission line). Feed structure  1914  is connected with support circuitry within card-shaped device  1902  (details not shown in FIG.  19 ). This configuration of antenna  1906  is particularly amenable for advantageous use with wireless communication devices connecting with personal computers (including laptop computers) via a PCMCIA structure. Card-shaped device  1902  may preferably be manufactured to ensure that its insertion to full insertion locus  1904  situates antenna  1906  sufficiently distant from any elements of a host device with which PCMCIA device  1900  is used to avoid undue interference from chassis structures or other RF interfering aspects of a host device. A representative such displacement in the case of a wireless device used with a laptop computer is on the order of one-fourth of the wavelength of the lowest frequency handled by the antenna (0.25λ L ). The ovoid dipole construction of antenna  1906  has a feed structure according to the teachings of the present invention and its construction is particularly amenable to wireless communications with a host device, such as a laptop computer, using impulse radio communications. Manufacture of antenna  1906  is easily accomplished by establishing radiating elements  1910 ,  1912  as planar ovoidal radiating elements on a flexible substrate, and then rolling the flexible substrate to form cylindrical substrate  1908 . 
     Spheroidal and ovoidal monopoles and dipoles have significant advantages over traditional antennas: 
     Compact size: at the low frequency limit, the span of the radiating elements is only about λ/6 to λ/10, often under half the size of the ˜λ/4 elements normally expected to be required for efficient radiation. This small size also helps the antenna to be non-dispersive, and thus well suited for short pulse transmission. 
     Well matched and efficient: spheroidal and ovoidal antennas can easily achieve VSWR&#39;s on the order of 2:1 or better. Some designs exhibit VSWR&#39;s as low as 1.2:1 across much of the band. This means that antennas have an efficiency of at least 90% to as high as 99%. These excellent matches over a large bandwidth mean that reflections (and hence, ringing) are minimized, despite the fact that no resistive loading is employed. These values are all matched to a 50Ω system, eliminating the need for an expensive broadband balun transformer. 
     Broadband: spheroidal and ovoidal antennas have fractional bandwidths of as much as 120%, i.e. covering two octaves. Fractional bandwidth is defined: 
         BW   ⁢   %     =         BW     3   ⁢   dB         f   C       =         f   U     -     f   L           1   2     ⁢     (       f   U     +     f   L       )               
 
where BW 3dB  is the 3 dB bandwidth, f C  is the center frequency, f U  is the upper frequency, and f L  is the lower frequency.
 
     Omni-directional: 
     spheroidal and ovoidal antennas have omni-directional patterns. 
     Ease of Manufacture: 
     a spheroidal monopole may be manufactured on a flat ground plane making it much easier to manufacture than a volcano smoke antenna or other spheroidal antenna. While both spheroidal dipoles and monopoles may be constructed from wire frame or plate type elements, ovoidal antennas are even more straightforward and simple in their manufacture. 
     Stable Gain: 
     Unlike most ultra wideband (UWB) antennas which exhibit an increasing gain with frequency, the gain of spheroidal and ovoidal antennas is remarkably stable across the performance band. 
       FIG. 20  is a table summarizing performance of various antennas vis-à-vis criteria considered important for a commercially successful impulse radio communication system. In  FIG. 20 , a variety of antennas are listed in the leftmost column. Arrayed for each antenna listed in the leftmost column are “YES” or “NO” comments for each of the following criteria: well-matched (i.e., to 50 ohms), efficient, non-dispersive, omni-directional, easy to make and small (i.e., compact). By inspection of  FIG. 20  one can observe that the CEO (i.e., circular, elliptical, oval) ovoidal planar antennas are the only antennas that indicate a “YES” for all categories. Spheroidal antennas are the only antennas that indicate only one “NO” among the important criteria indicated. 
     The favored antennas for impulse radio communication applications are ovoidal or spheroidal antennas that are fed at a “blunt” feed juncture having curvature in the area of feed junction, as described hereinbefore in detail, especially in connection with  FIGS. 14-19 . The feed structure employed may be coaxial cable, transmission line, twisted pair or other configurations of feed structure. The inventor has learned that it matters little whether the antenna is three-dimensional (i.e., spheroidal) or planar (i.e., CEO, or ovoidal). Moreover, it matters little whether the antenna shape is wholly “filled in” or merely outlines or circumscribes the intended shape of the embodiment employed by using mesh structure, plurality of plates (parallel or intersecting), or another approximation of the desired antenna shape. Radio communication performance varies somewhat among the various preferred embodiments, but not appreciably; other considerations than RF performance may dictate which embodiment to employ. Such other considerations may include cost, ease of manufacture, size, weight, robustness, aesthetics or other non-RF performance factors. 
     The most preferred embodiment is a planar elliptical dipole with elements aligned along the semi-minor axes and with about a 3:2 ratio between the semi-major and semi-minor axes. 
     Because the antennas disclosed in the present invention are capable of radiating very short, non-time-dispersive pulses, they are ideal for use in an array. Conventional elements in arrays exhibit undesirable grating lobes as later lobes of a pulse waveform interfere with earlier lobes. The antennas that are the subject of the present disclosure can emit short non-time-dispersive pulses that significantly mitigate the grating lobe problem. 
     Such short pulse waveforms allow the antennas of the present invention to be advantageously used in conjunction with corner, planar, convex cylindrical or concave cylindrical reflectors. When conventional antennas are used in a reflector, defocusing leads to undesired grating lobes. The short, non-time-dispersive pulses of the antennas of the present invention allow a reflected waveform to be defocused without leading to the grating lobes experienced when using conventional antennas. Defocusing a waveform without creating grating lobes permits higher gain and directivity than are achievable using prior art antenna elements. 
       FIG. 21  is a plan view of a schematic representation of a quadropole planar antenna according to the present invention. In  FIG. 21 , a quadropole antenna  2100  includes a first radiating element  2102 , a second radiating element  2104  and a third radiating element  2106 . A feed region  2110  includes connection points  2112 ,  2114  associated with first radiating element  2102 ; connection points  2114 ,  2114  associated with second radiating element  2104 ; and connection point  2120  associated with third radiating element  2106 . In an exemplary operational employment, in which antenna  2100  is, by way of example, connected with a coaxial cable, connection points  2112 ,  2114 ,  2116 ,  2118  may be connected with the grounding sheath of the coaxial cable (not shown in FIG.  21 ). In such an exemplary arrangement, connection point  2120  would be connected with the center pin of the coaxial cable (not shown in FIG.  21 ). Antenna  2100  thus has two gaps. Gap G 1  is established between ground pin connections  2112 ,  2114  and center pin connection  2120 . Gap G 2  is established between ground pin connections  2116 ,  2118  and center pin connection  2120 . Antenna  2100  is characterized by four beams of radiation in the plane of antenna  2100 . 
     Antennas constructed according to the teachings of the present invention have been observed to exhibit stable radiation patterns over at least a 4:1 bandwidth. That is, such antennas have a fractional bandwidth of at least 120%. Larger bandwidths are possible, but the radiation pattern of antennas having larger bandwidths will change at higher frequencies. For example, a dipole antenna radiation pattern within a particular bandwidth may shift to radiating in a quadropole antenna pattern at frequencies higher than the upper limit of the particular bandwidth. 
     It is to be understood that, while the detailed drawings and specific examples given describe preferred embodiments of the invention, they are for the purpose of illustration only, that the apparatus and method of the invention are not limited to the precise details and conditions disclosed and that various changes may be made therein without departing from the spirit of the invention which is defined by the following claims: