Abstract:
A non-coherent method for (PN) code synchronization and eliminates the need for prior clock synchronization between a transmitter and receiver. The method further eliminates any need for prior knowledge of a transmitted data sequence to provide synchronization. In the method, based on a coarse estimation of the carrier frequency, a received CDMA intermediate frequency (IF) signal is baseband converted to in-phase (I) and quadrature-phase (Q) components. The baseband I and Q components are then used to calculate an estimated delay {circumflex over (τ)} at a PN synchronization roll-over point. The estimated delay {circumflex over (τ)} is used in an intermediate data sequence in the synchronization roll-over calculation process to more accurately estimate carrier frequency error and phase offset for the IF carrier signal used in the baseband conversion process. The estimated carrier signal is then provided as feed back to correct for the coarse estimate of the carrier frequency error and phase offset, thus adaptively correcting for Doppler shift in a received signal.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to a method for performing delay. estimation and carrier synchronization on quadrature phase shift keying (QPSK) or offset QPSK (OQPSK) modulated code division multiple access (CDMA) systems. More particularly, the present invention provides a method for acquisition of a pseudo-noise (PN) synchronization point as well as Doppler shift in a received carrier signal. 
     2. Description of the Related Art 
     The motion of a receiver relative to a transmitter, such as a cell phone relative to a cell site, can generate a Doppler frequency and phase shift in a carrier signal. Such a frequency or phase shift can cause an increase in the bit error rate for synchronized users. 
     CDMA technology focuses on artificially increasing the bandwidth of a signal as a method of spread spectrum. Bandwidth is increased by breaking each bit into a number of sub-bits called“chips”. Assuming each bit is broken into 10 chips, the result is an increase in data rate by 10. By increasing the data rate by 10, bandwidth is also increased by 10. 
     Each original CDMA bit is divided into chips by multiplying the bit by a pseudo-noise (PN) code. A PN-code is an arbitrary sequence typically ranging between −1 and 1. Multiplying each of the original modulated signal bits by the PN-code results in the original bits being dividing into smaller chips, hence, increasing bandwidth. The-greater number of chips which the PN-code creates results in a wider bandwidth proportional to the number of chips. 
     To create a transmitted signal with PN-coding, a transmitter first modulates a message at a higher carrier frequency. For spread spectrum, all messages are modulated on the same carrier. After modulation, each signal is then multiplied by a PN-code in the transmitter to spread the bandwidth. Multiplying by the PN-code to spread signal bandwidth can also be done before carrier modulation. 
     In the receiver, the incoming signal is the spread spectrum signal. In order for the receiver to extract a transmitted message, the incoming signal is multiplied by the same PN-code used in the transmitter. With the PN-code ranging between −1 and 1, multiplying by the same PN-code in the receiver effectively cancels out the PN-code on the particular message. 
     With a Doppler shift due to the receiver moving relative to the transmitter, cancellation of the Doppler shift must occur to establish synchronization. Synchronization can be divided into two steps: acquisition and tracking. Acquisition is the process of roughly aligning the PN-code of the transmitted signal with the identical PN-code of the received signal. Tracking occurs after acquisition, and maintains a tight alignment of the two PN-codes over time. Misalignment of the transmit and receive PN-codes results in noise being generated in the received signal. The more severe the misalignment, the greater the bit error rate can be for the received signal. 
     The acquisition process for synchronization in the past required prior clock synchronization, including both frequency and phase synchronization, between the transmitter and receiver, or a prior knowledge of the transmitted data sequence. 
     SUMMARY OF THE INVENTION 
     In accordance with the present invention a non-coherent method for pseudo-noise (PN) code synchronization is provided eliminating the need for clock synchronization between a transmitter and receiver, and further eliminating a need for prior knowledge of the transmitted data sequence. The method further performs carrier synchronization simultaneously with PN synchronization. 
     In the method, based on a coarse estimation of carrier frequency error and phase offset, the received CDMA intermediate frequency (IF) signal is baseband converted to in-phase (I) and quadrature-phase (Q) components. The baseband I and Q components are then used to calculate an estimated delay at a PN synchronization roll-over point for the carrier signal. Once the PN synchronization roll-over point is determined, the delay calculated is used in an intermediate data sequence in the synchronization roll-over calculation process to more accurately estimate carrier frequency and phase delay for the carrier signal. The estimated carrier signal is then provided as feed back to correct for the coarse estimate of the carrier frequency error and phase offset, thus adaptively correcting for Doppler shift in the received signal. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     Further details of the present invention are explained with the help of the attached drawings in which: 
     FIG. 1 shows a high level block diagram for components used to provide carrier synchronization according to the present invention; and 
     FIG. 2 shows a detailed block diagram for the PN synchronization circuit of FIG.  1 . 
    
    
     DETAILED DESCRIPTION 
     I. Overview 
     FIG. 1 shows a high level block diagram for components used to provide carrier synchronization in accordance with the present invention. As shown, a CDMA RF signal is received and down converted to an intermediate frequency (IF) signal with a frequency ω IF  and phase φ IF  in downconverter  100 . The IF signal is then baseband converted to provide I and Q components with a coarse estimation of the carrier frequency error and phase offset in baseband (BB) converter  102 . The I and Q components from the baseband converter  102  are then used in the PN synchronization circuit  104  to determine delay estimates using formulas according to the present invention. 
     Signals from the PN synchronization circuit  104  are provided to peak detector  106 , and the output of the peak detector  106  provides an estimate of the delay in the received signal {circumflex over (τ)} at the PN spreading sequence roll-over point. Smaller search windows can then be used in the calculations made by the PN synchronizer circuit  104  with results provided to the peak detector  106  until an estimate of the delay {circumflex over (τ)} is obtained at the PN spreading sequence roll-over point with a desired resolution. 
     Once a desired PN synchronization is obtained from the output of the peak detector  106 , the delay {circumflex over (τ)} calculated at the roll-over point is used in the PN synchronizer circuit  104  and an intermediate data sequence from the PN synchronizer circuit  104  is provided to the frequency and phase delay estimator  108 . The frequency and phase estimator  108  then provides a more accurate estimate for frequency and phase delay ( 2 Δω, 2 Δφ) for the carrier signal. The frequency and phase delay from estimator  108  is provided to an IF frequency and phase adjuster  110  which generates a feed back signal with a frequency ω′ IF =ω IF +Δω and phase φ′ IF =φ IF +Δφ to the baseband converter  102  to provide a better coarse estimate of the carrier frequency error and phase offset for baseband conversion. After several iterations, the circuit of FIG. 1 can provide a very accurate indication of the carrier delay, thus adaptively correcting for Doppler shift in the received signal. 
     II. Delay Estimation 
     The baseband converted signal from baseband converter  102 , as provided to the PN synchronization circuit  104 , can be mathematically modeled in the following way:                  R   rcv          (   t   )       =       R        (   t   )                         (       Δ                 ω                 t     +     Δ                 φ       )                   (   1   )                            =       I        (   t   )       +                        Q        (   t   )                     (   2   )                                
     where, R(t) is the received baseband signal with perfect carrier synchronization, and Δω and Δφ are the coarse carrier frequency error and phase offset for the baseband I and Q signal. The baseband converted in-phase and quadrature-phase component signals are represented by I(t) and Q(t) respectively. 
     In the PN synchronization circuit  104 , the in-phase and quadrature-phase components (I(t) and Q(t)) and the π/4 phase shifted in-phase and quadrature-phase components (I′(t) and Q′(t)) are used in a formula to determine delay with respect to a short PN spreading sequence roll-over point. For the formula, I′(t) and Q′(t) can be represented as follows:                  R   rcv   ′          (   t   )       =         R   rcv          (   t   )                         (     π   /   4     )                   (   3   )                            =         I   ′          (   t   )       +                          Q   ′          (   t   )                     (   4   )                            =     1   /       2          [       (       I        (   t   )       -     Q        (   t   )         )     +             (       I        (   t   )       +     Q        (   t   )         )         ]                   (   5   )                                
     An estimated delay, {circumflex over (τ)} with respect to the in-phase (I) and quadrature-phase (Q) short PN spreading sequence roll-over point can be expressed as:                τ   ^     =       max     τ   ∈     [       τ   b     ,     τ   e       ]              [         (       ∑     η   =   0       N   -   1            x        (     t   +   τ     )         )     2     +       (       ∑     η   =   0       N   -   1              x   ′          (     t   +   τ     )         )     2       ]               (   6   )                                
     where,                x        (     t   +   τ     )       =       I        (     t   +   τ     )            Q        (     t   +   τ     )              I   PN          (   t   )              Q   PN          (   t   )                 (   7   )                   x   ′          (     t   +   τ     )       =         I   ′          (     t   +   τ     )              Q   ′          (     t   +   τ     )              I   PN          (   t   )              Q   PN          (   t   )                 (   8   )                            =       1   /   2          (         I   2          (   t   )       -         Q   2          (   t   )              I   PN          (   t   )              Q   PN          (   t   )                         (   9   )                                
     t is ηT c , where η=0,1,2, . . . N−1 indicating the signals X(t+τ) and x′(t+τ) are sampled at a rate of 1/T e  in equation (6) 
     T c  is a duration of a PN chip 
     τb is the beginning of a search window for delay estimation 
     τ e  is the end of a search window for delay estimation 
     N is the total number of samples used in the estimation process 
     I PN (t) is the in-phase short PN spreading sequence 
     Q PN (t) is the quadrature-phase short PN spreading sequence 
     FIG. 2 shows a detailed block diagram for the PN synchronization circuit  104  of FIG. 1 with components used to determine x(t+τ) and x′(t+τ) according to equations (7) and (8). FIG. 2 also shows connection of components of the PN synchronization circuit  104  to the peak detector  106  to enable determination of {circumflex over (τ)} of equation (6). FIG. 2 further shows connection of the PN synchronization circuit  104  to the frequency and phase estimator  108  and frequency and phase adjuster  110 . 
     As shown in FIG. 2, I(t) is received by a variable delay circuit  200 . The variable delay circuit  200  can be controlled to provide a desired delay τ enabling the output of the delay circuit  200  to provide the component I(t+τ) for equation (7) which can be used in equation (6). Q(t) is received by a variable delay circuit  202 . The variable delay circuit  202  is controlled to provide a desired delay T enabling the output of the delay circuit  202  to provide the component Q(t+τ) for equation (7) which can be used in equation (6). 
     The signals I′(t) and Q′(t) can be provided to the PN synchronization circuit  104  from baseband converter  102 . With I′(t) and Q′(t) provided, delay circuits  204  and  206  can be included in the PN synchronization circuit, as shown in FIG.  2 . As with the delay circuits  200  and  202 , the delay circuits  204  and  206  can be controlled to provide a desired delay τ so that the output of circuits  204  and  206  provide the components I′(t+τ) and Q′(t+τ) of equation (8) which can be used in equation (6). 
     Without components to provide the signals I′(t) and Q′(t), the PN synchronization circuit  104  might include circuitry to square the signals I(t) and Q(t), subtract the difference, and multiply the result by {fraction (1/2 )} to provide the component ½(I 2 (t)−Q 2 (t)) of equation (9) to solve for x′(t+τ) instead of solving for x′(t+τ) using equation (8). 
     The PN spreading sequences are known parameters, so the terms I PN (t) and Q PN (t) in any of equations (7), (8) and (9) can be pre-calculated. In the circuit of FIG. 2 to provide the result for the component x(t+τ) of equation (7), the value I PN (t) is multiplied with I(t+τ) in multiplier  210 . The value Q PN (t) is further multiplied with Q(t+τ) in multiplier  212 . Multiplier  220  then combines the outputs of multipliers  210  and  212  to complete the function I(t+τ)I PN (t)Q(t+τ)Q PN (t) of equation (7). 
     To provide the result for the component X′(t+τ) of equation (8) in the circuit of FIG. 2, the value I PN (t) is multiplied with the output of circuit  204  in multiplier  214 . The value Q PN (t) is further multiplied with the output of circuit  206  in multiplier  216 . Multiplier  222  then combines the outputs of multipliers  214  and  216  to complete the function I′(t+τ)I PN (t)Q′(t+τ)Q PN (t) of equation (8). 
     The output of multiplier  220  is provided to a summer  230  which sums the results from the output of multiplier  220  N times. The output of the summer  240  is provided to squaring circuit  240 . The output of the squaring circuit  240 , thus, provides the component          (       ∑     η   =   0       N   -   1            x        (     t   +   τ     )         )     2                          
     from equation (6). Similarly, the output of multiplier  222  is provided to summer  232  which sums the results from the output of multiplier  222  N times. The output of summer  232  is provided to squaring circuit  242 , so that the output of squaring circuit  242  provides the component          (       ∑     η   =   0       N   -   1              x   ′          (     t   +   τ     )         )     2                          
     of equation (6). The outputs of the summers  240  and  242  are provided to an adder  244 , and the output of the adder is provided to a peak detector  106 . The output of the peak detector  106  then provides the result for {circumflex over (τ)} of equation (6). 
     As can be seen from the components of the PN synchronization circuit of FIG. 2, although the components I PN (t) and Q PN (t) needed to solve equation (6) are known, the components I(t) and Q(t) must be interpolated in real time to determine the delay {circumflex over (τ)}. 
     For an initial interpolation of a value for {circumflex over (τ)} a wide search window between τ b  and τ e  can be used with a low resolution to estimate the delay. Once an estimate of the delay {circumflex over (τ)} with the initial search window is obtained, a much smaller search window can be applied in consecutive data frames. With a communication system having a slow time varying process, a smaller search window can be used while tracking of the acquired carrier is maintained. Once the desired PN synchronization rollover point is obtained with a desired search window and resolution, the frequency and phase estimator  108  is activated to determine carrier frequency error and phase offset. 
     III. Limit of Coarse Carrier Frequency Error 
     A large coarse carrier frequency error can have a detrimental effect on the PN synchronization delay estimation process. The formula of equation (6) can be used if the following condition is met: 
     
       
         Δf t ≦f s ·{fraction (1/16)}·1/N  (10) 
       
     
     where Δf t  is the carrier frequency error in the baseband signal and f s  is the sampling rate of the broadband I and Q signals. Typically the carrier frequency error can be initially estimated with an accuracy which enables the condition of equation (10) to be met using a phase locked loop (PLL) in the baseband converter. In CDMA systems a reasonable choice of N for equation (6) can be 64 samples. If the sampling rate of the baseband I and Q signal is 4×1.2288 MHz, then the maximum frequency error tolerated for equation (6) is ±({fraction (1/16)}·{fraction (1/64)}·4·1.2288 MHz) or ±4.8 kHz. If the frequency error Δf t  of the carrier signal is outside the range of equation (10), but falls within a higher frequency tolerance the formula of equation (6) can be replaced with the following equation:                τ   ^     =       max     τ   ∈     [       τ   b     ,     τ   e       ]              [         ∑     η   =   0       N   -   1              (       x   LP          (     t   +   τ     )       )     2       +       ∑     η   =   0       N   -   1              (       x   LP   ′          (     t   +   τ     )       )     2         ]               (   11   )                                
     where, 
     X LP (t+τ) is the output of the signal X(t+τ) to a low pass filter with a cutoff frequency at twice the frequency tolerance, and 
     X LP ′(t+τ) is the output of the signal X′(t+τ) to a low pass filter with a cutoff frequency at twice the frequency tolerance. 
     The remaining parameters of equation (11) are identified previously with respect to equation (6). To provide a value for T using equation (11), the circuitry of FIG. 2 can be modified by replacing the summer  230  and squaring circuit  240  with a low pass filter with a cutoff frequency at twice the frequency tolerance Δf t  followed by a total energy measuring element. The summer  232  and  242  are likewise replaced by a low pass filter and total energy measuring element. Existing algorithms for PN synchronization have similar limitations on frequency error as can be seen in Andrew Viterbi,  CDMA Principles of Spread Spectrum Communication , section 3.2.3, pp. 45-47, June 1995. 
     III. Carrier Synchronization 
     After the PN synchronization roll-over point is acquired within the error limits as described above, both carrier frequency error and phase offset can be estimated using the frequency and phase estimator  108 . The estimator  108  receives the output from the mixers  220  and  222  of FIG. 2 with the delay {circumflex over (τ)} provided at the output of peak detector  106  set in the variable delay devices  200 ,  202 ,  204  and  206 . The estimator  108  applies these received signals to the formulas described below to provide an estimate for the frequency error Δf and phase offset Δφ in the carrier signal. To derive the formula used in the estimator  108 , it can be shown that the signal: 
     
       
         s(t)=x(t+{circumflex over (τ)})+i·x′(t+{circumflex over (τ)})  (12) 
       
     
     can be modeled as a single complex sinusoid in additive noise. Moreover, the frequency and phase of the signal s(t) has twice the error of the coarse carrier frequency and phase. Therefore, the carrier signal can be synchronized very accurately using sinusoidal parameter estimators on the signal s(t). The maximum likelihood (ML) estimator for a single complex sinusoid in additive white Gaussian noise is the Periodogram method. Using the Periodogram method, carrier frequency error and phase offset can be estimated as follows:                2      Δ                 f     =       max     f   ∈     [         -   Δ                     f   t       ,     Δ                   f   t         ]                     ∑     η   =   0       M   -   1              s        (   t   )            exp        (       -   2                   π                 f                 t     )                          (   11   )                 2      Δ                 φ     =     arctan        [         ∑     η   =   0       M   -   1              s        (     t   ′     )          sin                 2        π        (     2                 Δ                 f     )            t   ′             ∑     η   =   0       M   -   1              s        (     t   ′     )          cos                 2        π        (     2      Δ                 f     )            t   ′           ]               (   12   )                                
     where, 
     t is ηT c , where η=0,1,2, . . . N−1, 
     T c  is a duration of a PN chip, 
     Δf t  is the frequency tolerance of the algorithm described in equation (10) 
     M is the total number of samples used in the carrier synchronization process. Note for better estimation, M can be a number much higher than N. 
     The circuit  108  then provides the value 2Δf according to equation (13) times 2Π, totaling 2Δω, to the frequency adjuster  110 . The circuit  108  further provides the value 2Δφ from equation (14) to the frequency adjuster  110 . The frequency adjuster  110  then provides a signal with a frequency ω′ IF  equal to the frequency of the carrier ω IF  as corrected for the error Δω as feedback to the broadband converter  102 . The frequency adjuster  110  signal further is provided with a phase φ′ IF  equal to the phase of the original carrier φ IF  as corrected for the phase offset Δφ. 
     Since equation (14) estimates twice the phase rather than absolute phase, the estimated phase can have a 180° phase ambiguity. In other words, the absolute phase change will be either the estimated phase offset Δφ from equation (14), or the phase offset from equation (14) plus 180°. 
     In most CDMA systems at the beginning of a channel a known data sequence is sent so that the receiver can acquire carrier synchronization. For example in an IS-95 CDMA wireless system, either a pilot channel, a preamble in an access channel, or a preamble in a traffic channel is sent from a transmitter to a receiver to assure successful handoff. If the method of the present invention is used to continuously synchronize to an incoming data sequence after synchronization using such a known data sequence, a 180° phase ambiguity will not be present. If the method of the present invention is used to demodulate an intermediate data sequence which has not been synchronized using a known data sequence, the 180° phase ambiguity can result in an inversion in the demodulated data sequence which can still be resolved by a channel decoder, such as a cyclical redundance check (CRC), or orthogonal demodulation if present in the system. 
     Although the present invention has been described above with particularity, this was merely to teach one of ordinary skill in the art how to make and use the invention. Many additional modifications will fall within the scope of the invention, as that scope is defined by the claims which follow.