Abstract:
A method and apparatus for use with an induction machine system including a controller and d and q-axis current feedback loops, the controller receiving a frequency command signal and generating d and q-axis voltage command signals, the method for limiting load current to a level below a limit current at low operating frequencies, the method comprising the steps of identifying an operating frequency as a function of the d and q-axis feedback currents, where the operating frequency is below a low threshold value: comparing a feedback current to the limit current; and where the feedback current exceeds the limit current, reducing the q-axis voltage command value.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     Not applicable. 
     STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT 
     Not applicable. 
     BACKGROUND OF THE INVENTION 
     The field of the invention is motor controllers and more specifically a method and apparatus for limiting current in an open loop adjustable frequency motor drive at low operating frequencies. 
     Induction motors have broad application in industry. An induction motor system typically includes a driver or controller, a power conversion configuration and an induction motor itself. The power conversion configuration generally receives power via supply lines and converts the received power into a form that can be provided to the motor thereby causing a motor rotor to rotate. The conversion configuration typically includes a plurality of semiconductor switching devices that link the supply lines to motor terminals and, based on switch turn on and turn off cycles, provide power to the motor phases linked thereto. 
     One common type of motor is a three-phase induction motor that includes a stator and a rotor. The stator typically forms a cylindrical stator cavity. One common rotor design includes a “squirrel cage winding” in which axial conductive rotor bars are connected at either end by shorting rings to form a generally cylindrical structure. The rotor is mounted in the stator cavity for rotation about a rotor axis. The stator windings are linked to three separate phases of the converter configuration to receive currents therefrom. The stator currents are controlled so that their combined effect is to generate a magnetic stator field that rotates about the stator cavity. The rotating stator field flux cuts across the conductive rotor bars and induces (hence the label “inductance motor”) cyclic current flows through the bars and across the shorting rings. The cyclic rotor bar current flows in turn produce a rotor field. Interaction (e.g., pulling or pushing action) between the rotor field and the stator field causes the rotor to rotate. 
     By using induced rotor current to generate the rotor field, the need for slip rings or brushes (i.e., wearable mechanical components) is eliminated which renders induction type motors relatively maintenance-free and reduces overall costs associated with motor design. Among other reasons, relatively limited costs have made inductance motors preferred for many applications throughout industry. 
     To a first approximation the torque (i.e., rotational force on the rotor) and speed of an induction motor may be controlled by changing the frequency of the driving voltage and thus the angular rate of the rotating stator field. Generally, for a given torque, increasing the stator field rate will increase the rotor speed (which generally follows the stator field). Alternatively, for a given rotor speed, increasing the frequency of the stator field will increase the torque by increasing the slip, that is, the difference in speed between the rotor and the stator field. An increase in slip increases the rate at which flux lines are cut by the rotor bars thereby increasing the rotor-generated field and thus the force or torque between the rotor field and stator field. 
     Referring to FIG. 10, the rotating phasor  13  of the stator magneto motive force (“mmf”) will generally form some angle α with respect to the phasor of rotor flux  19 . The torque generated by the motor is proportional to the magnitudes of these phasors  13  and  19  but also is a function of their angle α. The maximum torque is produced when phasors  14  and  18  are at right angles to each other (e.g., α=90°) whereas zero torque is produced if these phasors are aligned (e.g., α=0°). Phasor  13  may, therefore, be usefully decomposed into a torque producing component  15  perpendicular to the phasor  19  and a flux component  17  parallel to rotor flux phasor  18 . 
     These two components  15  and  17  of the stator mmf are proportional, respectively, to two stator currents i qe , a torque producing current, and i de , a flux producing current, which may be represented by orthogonal vectors in a rotating or synchronous reference frame of the stator flux having slowly varying magnitudes. The subscript “e” is used herein to indicate that a particular quantity is in the rotating or synchronous frame of stator flux. 
     Accordingly, in controlling an induction motor, it is generally desired to control not only the frequency of the applied voltage (hence the speed of the rotation of the stator flux phasor  13 ) but also the phase of the applied voltage relative to the current flow and hence the division of the currents through the stator windings into the i qe  and i de  components. Control strategies that attempt to independently control currents i qe  and i de  are generally termed field oriented control (FOC) strategies. 
     The production of any given set of currents i qe  and i de  requires that the stator be excited with voltages V qe  and V de  as follows: 
     
       
           V   qe =( R   s )( i   qe )+(2 πf   e )(λ rated )  Eq. 1 
       
     
     
       
           V   de =( R   s )( i   de )  Eq. 2 
       
     
     where 
     R s =stator resistance; 
     i qe , i de =synchronous motor currents aligned with the d and q-axis typically reflecting motor load and no load currents, respectively; 
     f e =electrical field frequency in Hertz; and 
     λ rated =stator flux linkage=motor nameplate voltage/motor nameplate frequency (in Hertz). 
     The first terms on the right hand sides of each of Equations 1 and 2 are referred to as the stator resistive voltage drops. As the labels imply, the resistive voltage drops R s i qe  and R s i de  correspond to components of the voltage provided at a stator winding terminal that are dissipated by the stator winding resistance R s . Because the resistive drops are provided to boost the commanded voltages and, in effect, overcome the stator resistance R s , the resistive drops are often referred to as “voltage boost” terms. The second term 2πf e λ rated  on the right hand side of Equation 1 is referred to generally as a reactive voltage drop and, as its label implies, corresponds to the component of the voltage provided at the stator winding terminal that causes inductance or interaction between the stator and the rotor. 
     Equations 1 and 2 above are the fundamental command equations employed by most voltage/frequency controllers. To implement Equations 1 and 2, the controller has to be provided with several of the terms in each of Equations 1 and 2. 
     In order to minimize costs, often controller/converter configurations are designed to be useable for many different purposes (i.e., to drive many different load types). For instance, one controller/converter configuration may be capable of driving any of several differently sized three phase motors where the motors have different operating characteristics. Thus, when designing controller/converters, manufacturers typically do not know exact characteristics of loads that will be linked to and driven by the controller/converters and, therefore, some controller operating parameters have to be set by customers after system configuration is completed. 
     The rated flux λ rated  can be determined by dividing a name plate motor voltage by a nameplate frequency values while the stator winding resistance R s  is typically determined by performing a commissioning procedure (e.g., see U.S. Pat. No. 5,502,360 for a commissioning procedure to determine R s ). The d-axis current i de  may be determined in any of several different ways including use of a look-up table that correlates d-axis current with various motor parameters or by performing some type of commissioning procedure. Each of the d-axis current i de , the stator resistance R s  and the rated flux λ rated  are stored in a controller memory for use during motor operation. The d-axis current i de  typically is not adjusted during motor operation and therefore the d-axis voltage V de  is set upon commissioning. 
     In addition to the components described above, most controllers also include some type of feedback mechanism to ensure that an associated load (e.g., motor) operates in a desired fashion. To this end, typical feedback loops include a rotor speed feedback and d and q-axis current feedbacks. The feedback signals are generally compared to commanded signals and, where errors occur, the commanded signals are modified to force the load toward desired operating characteristics. For instance, where a feedback rotor speed is less than a commanded rotor speed, the rotor speed error can be used to command a higher electrical frequency thereby increasing slip and torque on the rotor and causing the rotor speed to increase by a percentage of the increase in the electrical frequency. 
     To implement Equations 1 and 2, after rated flux λ rated , stator resistance R s  and d-axis current i de  values are identified and stored in the motor controller memory, the controller receives a rotor speed command that indicates a desired motor rotor rotational speed. In addition, d and q-axis feedback currents i def  and i qef  are provided to the controller. The controller uses the commanded frequency and the feedback currents to generate suitable d and q-axis voltages V de  and V qe , respectively by solving Equations 1 and 2 above. Thereafter, the controller converts the d and q-axis voltages V de  and V qe  into three phase voltage commands to drive converter switches. 
     As with all electronic components, the switching devices that comprise the converter configuration are designed to operate within specific rated current operating ranges and will be damaged or may operate in unintended ways when driven outside the rated current ranges. Unfortunately, during induction motor operation, conditions have been known to occur that cause controllers to demand current levels outside rated ranges. For instance, when a load is increased, the load will generally slow the rotation of a motor rotor which causes a difference between a commanded frequency and an actual frequency. The frequency difference or error causes the controller to step up the commanded voltage thereby, referring again to FIG. 1, increasing the q-axis current i qe . At high frequencies where the reactive drop is ten or more times the resistive drop, a reactive drop adjustment (e.g., f e  adjustment) appreciably affects commanded voltage V qe  while at a low frequency where the reactive drop may be one-fifth or less of the resistive drop, a reactive drop adjustment may not be capable of avoiding a current trip. At some point, as the load is increased, the q-axis current i qe  exceeds the high end of a rated switch current range and switch damage or malfunction may occur. 
     To avoid switch damage/malfunction, most controllers now include a “current tripping” function wherein, when measured switch currents exceed the high end of a rated switch range, the control system trips and, in effect, cuts off current to the converter and load thereby protecting the converter switching devices. While tripping is clearly preferred to switch damage, tripping hinders system productivity and is to be avoided whenever possible. 
     To minimize current tripping, most controllers now include some type of current limiting feature. One common current limiting scheme reduces the commanded electrical frequency f e  when the upper end of the rated switch current range is exceeded. Referring again to Equation 1, when frequency f e  is reduced, the commanded q-axis voltage V qe  is reduced which in turn reduces the resulting q-axis current i qe . 
     Frequency reducing schemes work well at relatively high frequencies and poorly at low frequencies. This frequency based effectiveness difference is due to the fact that the commanded voltage splits between the resistive drop component R s i qe  and the reactive component 2πf e λ rated  and the ratio of resistive to reactive drops is in great part based on frequency f e . For example, at high frequencies (e.g., a name plate frequency) reactive drop component 2πf e λ rated  may be ten or more times resistive drop component R s i qe  while at low frequencies the reactive drop may be one-fifth or less of the resistive drop. At high frequencies where the reactive drop is ten or more times the resistive drop, a reactive drop adjustment (e.g., f e  adjustment) appreciably affects commanded voltage V qe  while at a low frequency where the reactive drop may be a fraction of the resistive drop, a reactive drop adjustment may not be capable of avoiding a current trip. Other sources of error that can cause positive current feedback are also contemplated including imperfect switching characteristics that result in unexpected winding current levels, reflected waves caused by long power supply cables, etc. 
     Prior known solutions to the current tripping problem at low operating frequencies simply stepped the commanded voltage V qe  to some level less than the voltage boost level R s i qe  and therefore resulted in sudden, unintended and undesirable changes in output torque to the load. 
     Thus, there is a need for an inexpensive method and/or apparatus that can smoothly control system current levels so as to avoid current trip conditions without causing undesirable torque pulsations at low operating frequencies. 
     BRIEF SUMMARY OF THE INVENTION 
     It has been recognized that the low frequency current tripping problem described above can be overcome by providing a voltage boost limiting mechanism that will hold the voltage boost level below a level that will cause a current limit condition. By implementing such a limiting scheme in conjunction with a frequency based current limiting scheme at higher frequencies, virtually all current tripping conditions, independent of frequency and independent of the source of excessive current, can be eliminated and overall smother system operation results. 
     To this end, the invention includes a method for use with an induction machine system including a controller and d and q-axis current feedback loops, the controller receiving a frequency command signal and generating d and q-axis voltage command signals, the method for limiting load current to a level below a limit current at low operating frequencies. The method comprises the steps of identifying an operating frequency, where the operating frequency is below a low threshold value: comparing a feedback current to the limit current and where the feedback current exceeds the limit current, reducing the q-axis voltage command value. 
     In one embodiment the step of comparing a feedback current includes comparing a q-axis feedback current to a maximum q-axis feedback current. In a more specific embodiment the method further includes the step of mathematically combining the limit current and a d-axis feedback current to identify the maximum q-axis feedback current. Here the step of mathematically combining may include taking the square root of the difference of the squares of the limit current and the d-axis current. Furthermore, the step of comparing a q-axis feedback current to a maximum q-axis feedback current may include subtracting the absolute value of the feedback current from the maximum q-axis feedback current to generate a difference value and the step of reducing includes reducing the q-axis voltage command value when the difference value is negative. Even more specifically, the controller may generate a nominal voltage boost value by multiplying a stator resistance value and the q-axis feedback current and the step of reducing may include multiplying the sign of the q-axis feedback current and the difference value to generate a signed difference value and mathematically combining the signed difference value and the nominal boost voltage to generate a limited boost voltage. Here the step of limiting further may include the step of mathematically combining the operating frequency and the limited boost value to generate the q-axis voltage command value. The threshold value may be less than one percent of a nameplate frequency for the induction machine. 
     In another embodiment the step of comparing a feedback current includes mathematically combining the d-axis and q-axis feedback currents to generate an instantaneous stator current and wherein the step of comparing includes comparing the instantaneous stator current to the limit current. Here, the step of mathematically combining may include taking the square root of the sum of the squares of the d-axis current and the q-axis current to generate the instantaneous stator current. Still more specifically, the step of comparing may include subtracting the instantaneous stator current from the current limit to generate a difference value and wherein the step of reducing includes reducing the q-axis voltage command value when the difference value is negative. Here a nominal voltage boost may be provided by a controller user and the step of reducing may include mathematically combining the difference value and the nominal boost voltage to generate a limited boost voltage. More specifically the step of limiting may further include the step of mathematically combining the operating frequency and the limited boost value to generate the q-axis voltage command value. 
     The invention also includes a method for use with an induction machine system including a controller and d and q-axis current feedback loops, the controller receiving a frequency command signal and generating d and q-axis voltage command signals, the method for limiting load current to a level below a limit current at low operating frequencies. Here the method comprises the steps of identifying an operating frequency, where the operating frequency is below a low threshold value: mathematically combining the d-axis feedback current and the limit current to generate a maximum q-axis feedback current; comparing the q-axis feedback current the maximum q-axis current and, where the q-axis feedback current exceeds the maximum q-axis current, reducing the q-axis voltage command value. In one aspect the step of mathematically combining includes taking the square root of the difference of the squares of the limit current and the d-axis current. 
     The invention also includes an apparatus for use with an induction machine system including a controller and d and q-axis current feedback loops, the controller receiving a frequency command signal and generating d and q-axis voltage command signals, the apparatus for limiting load current to a level below a limit current at low operating frequencies. The apparatus comprises a processor running a pulse sequencing program to perform the steps of: identifying an operating frequency, where the operating frequency is below a low threshold value, comparing a feedback current to the limit current and where the feedback current exceeds the limit current, reducing the q-axis voltage command value. 
    
    
     These and other objects, advantages and aspects of the invention will become apparent from the following description. In the description, reference is made to the accompanying drawings which form a part hereof, and in which there is shown a preferred embodiment of the invention. Such embodiment does not necessarily represent the full scope of the invention and reference is made therefore, to the claims herein for interpreting the scope of the invention. 
     BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS 
     FIG. 1 is a schematic diagram of an exemplary motor control system according to the present invention; 
     FIG. 2 is a schematic diagram illustrating the controller of FIG. 1 in greater detail; 
     FIG. 3 is a schematic diagram illustrating the d-axis voltage reference generator of FIG. 2 in greater detail; 
     FIG. 4 is a schematic diagram illustrating the nominal torque boost of FIG. 2 in greater detail; 
     FIG. 5 is a schematic diagram illustrating the frequency based current limiter of FIG. 2 in greater detail; 
     FIG. 6 is a schematic diagram illustrating the torque boos limiter of FIG. 2 in greater detail; 
     FIG. 7 is a graph illustrating two motor currents generated without the low frequency current limiting method of the present invention; 
     FIG. 8 is similar to FIG. 7 albeit illustrating two currents where the inventive method has been adopted; 
     FIG. 9 is a simple voltage/frequency voltage reference generation curve; 
     FIG. 10 is a schematic diagram illustrating various motor operating parameters; and 
     FIG. 11 is a schematic diagram illustrating a second embodiment of the q-axis voltage reference generator of FIG. 2; and 
     FIG. 12 is a flow chart illustrating an inventive method performed by the controller of FIG.  1 . 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     A. Hardware 
     In the description that follows, an “e” subscript is used to denote signals/values in a synchronous or rotating (as opposed to stationary) frame of reference, an “f” subscript denotes a feedback signal, a “q” subscript denotes a q-axis value, a “d” subscript denotes a d-axis value, “u”, “v” and “w” subscripts denote signals corresponding to the three phases of the control system, an “*” superscript denotes a commanded value, a “lim” subscript denotes a limit value, “i” and “p” subscripts denote integral and proportional values, a “rated” subscript denotes a rated value, an “s” subscript denotes a stator related signal. 
     Referring now to the drawings wherein like reference numerals correspond to similar elements throughout the several views and, more specifically, referring to FIG. 1, the present invention will be described in the context of an exemplary motor control system  10  including a user interface  12 , a controller  14 , an AC voltage source  16 , a converter configuration  18 , a three-phase motor  20 , a three-to-two phase and stationary to synchronous transformer  22  and various supply and control lines which will be described in more detail below. User interface  12  is used to provide various system operating characteristics including a stator resistance value R s , a current limit value i lim , a rated flux value λ rated , a proportional gain factor K p , an integral gain factor K i , an acceleration/deceleration gain factor K ad , a minimum electrical frequency f elow  and a maximum electrical frequency f ehigh , a d-axis current value i dec  and a command frequency f*. 
     The acceleration/deceleration gain factor K ad  is a value that is typically user selectable and will depend on the type of load being driven by the system  10 —as its label implies, this rate simply indicates how quickly the load should be accelerated and decelerated. The minimum and maximum frequencies f elow  and f ehigh  are similarly user selectable and will typically be determined as a function of load characteristics (e.g., with certain loads the user will want to limit rotational frequencies to within a specific operating range). The current limit i lim  is typically a rated or name plate current value which indicates an optimal maximum current corresponding to the converter configuration  18 . The K p  and K i  values are simply scalars that are used to adjust how quickly the system  10  adjusts to reduce operating errors. The resistance R s  is determined via a commissioning procedure, d-axis current value i dec  is determined either via a commissioning procedure or via some suitable look-up table and the rated flux λ rated  is determined by dividing a motor name plate voltage value by a nameplate frequency (in Hz). The command frequency f* indicates a desired rotor speed. 
     Controller  14  use the received signals from interface  12  along with various feedback signals to generate three-phase voltage command signals V v *, V u * and V w * on trigger lines  26 . The trigger lines  26  are linked to converter configuration  18  which, as well known in the art, includes both an AC/DC converter and an inverter. AC voltage source  16  provides three-phase AC voltage to the converter configuration  18  which converts that AC voltage to a DC voltage and then converts the DC voltage to three-phase AC voltage on three motor supply lines  28 . 
     Supply lines  28  are each separately linked to one of the three phases of motor  20  to provide voltages V v , V u  and V w  thereto, respectively. The phase voltages cause currents within the stator windings of motor  20  that, together, generate a rotating stator flux field within a stator cavity (not illustrated). A motor rotor is mounted within the stator cavity for rotation about a rotation axis. The rotating stator flux field induces currents in the motor rotor bars which in turn generate a rotor flux field that interacts with the stator flux field to cause the rotor to rotate within the stator cavity. 
     Referring still to FIG. 1, two separate currents sensors (not illustrated) are linked to the supply lines corresponding to the u and w motor phases to provide two-phase current feedback signals i uf  and i wf  in the stationary frame of reference to transformer  22  via lines  30  and  32 . As well known in the controls art, transformer  22  converts the stationary two-phase currents i uf  and i wf  to two-phase synchronous feedback currents i qef  and i def  on lines  34  and  36 , respectively. Feedback currents i qef  and i def  are provided to controller  14 . 
     Referring now to FIG. 2, controller  14  is illustrated in greater detail and includes a d-axis voltage reference generator  40 , a q-axis voltage reference generator  42 , a frequency based current limiter  48  and a synchronous to stationary and two-to-three phase transformer  50 . Q-axis voltage reference generator  42  includes both a nominal torque boost  44  and a torque boost limiter  46 . The synchronous to stationary and two-to-three phase transformer  50  is well known in the art and therefore will not be explained here in detail. It should suffice to say that transformer  50  receives synchronous d and q-axis voltage command signals V de  and V qe , respectively, and transforms those two synchronous voltages to two-phase stationary voltages and then transforms the two-phase stationary voltages into three-phase stationary command voltages V v *, V u *, and V w * on trigger lines  26  which are linked to the converter configuration switches (e.g., converter  18  as illustrated in FIG.  1 ). The d-axis voltage reference generator  40 , nominal torque boost  44 , frequency based current limiter  48  and torque boost limiter  46  are illustrated in greater detail in FIGS. 3,  4 ,  5  and  6 , respectively. 
     Referring now to FIG. 3, d-axis voltage reference generator  40  includes a single multiplier  52  and, consistent with Equation 2 above, multiplies the stator resistance value R s  and the d-axis synchronous current value i dec  to generate the synchronous d-axis voltage value V de  which is provided to transformer  50  (see again FIG.  2 ). 
     Referring to FIG. 4, the nominal torque boost  44  includes a single multiplier  54  that multiplies the stator resistance R s  and the synchronous q-axis feedback current i qef  to generate the nominal voltage boost or resistive voltage drop value R s i qef  that comprises the first term on the right-hand side of Equation 1 above. This nominal value R s i qef  is provided to the torque boost limiter  46  as illustrated in FIG.  2 . 
     Referring now to FIG. 5, the frequency based current limiter  48  receives the q and d-axis feedback currents i qef  and i def , respectively, the current limit value i lim , the frequency command f e  and the high and low frequency limit values f ehigh  and f elow , respectively, and generates an output frequency value f eout  that is provided to torque boost limiter  46 . Current limiter  48  includes, in at least one embodiment, first and second square modules  60  and  62 , respectively, five summers including a first summer  49 , a second summer  66 , a third summer  84 , a fourth summer  86  and a fifth summer  81 , a single square root module  64 , two multipliers including a first multiplier  68  and a second multiplier  72 , three scalar modules including a proportional scalar module  70 , an integral scalar module  76  and an acceleration/deceleration scalar module  77 , a single sign module  74 , a digital integrator module  78 , a single pole switch  79 , a double pole switch  80 , a frequency range limiter  82  and a comparator  83 . First square module  60  receives the synchronous q-axis feedback current i qef  and, as its label implies, provides the square of the feedback current i qef . Similarly, second square module  62  received the synchronous d-axis feedback current i def  and squares that received value. The outputs of square modules  60  and  62  are added together by first summer  49  and the square root module  64  provided the square root of that sum as an output to second summer  66 . The output of square root module  64  is a stator current feedback value i sf  corresponding to the instantaneous stator current magnitude. 
     Referring still to FIG. 5, the stator feedback current i sf  is subtracted from the stator current limit value i lim  via second summer  66  and the difference between the two values Δi s  is provided to first multiplier  68 . Difference value Δi s  indicates whether or not the instantaneous motor current exceeds the limit value i lim . Here, where the difference value Δi s  is positive, the instantaneous current is less than the limit current i lim  and the current limiting scheme is not activated. Where the difference value Δi s  is negative, however, the instantaneous current exceeds the limit current i lim  and a frequency reducing scheme is activated to reduce system current. 
     Difference value Δi s  is provided to comparator  83  which compares difference value Δi s  to zero and controls switches  79  and  80  as a function of the comparison. When value Δi s  is positive (i.e., the current limit has not been exceeded) comparator  83  opens switch  79  and links switch  80  to a “no limit” input pole to cause normal system operation. When value Δi s  is negative (i.e., the current limit i lim  has been exceeded), comparator  83  closes switch  79  and links switch  80  to a second input pole to reduce the output frequency f eout  of module  48 . 
     Referring still to FIG. 5, second multiplier  72  receives the q-axis feedback signal i qef  along with an electrical frequency output signal f eout  generated by module  48  and multiplies those two signals to generate a signal having a sign that indicates whether or not the load is in a motoring state or a regenerating state and the direction of the load. Here, the output of multiplier  72  is provided to sign module  74  to identify the sign (i.e., + or −) of the received signal. The sign is provided to multiplier  68  which multiplies the sign by difference value Δi s . The output of first multiplier  68  is provided to each of the first and second scalar modules  70  and  76 , respectively. 
     Referring still to FIG. 5, scalar modules  70  and  76  multiply the signed difference value Δi s  by scalar gains K p  and K i , respectively, and provide the stepped up values as current limit values to the current limit inputs of switches  79  and  80 , respectively. As indicated above, the K i  and K p  values determine how quickly the system forces system frequency lower when an excessive current condition occurs and are, at least in some embodiments, user selectable. 
     Output frequency f eout  is provided to summer  81  which subtracts output frequency f eout  from command frequency f* to generate a frequency error value Δf. Error value Δf is stepped up by the acceleration/deceleration gain K ad  at block  77  and the stepped up error value K ad Δf is provided as the “no current limit” input to the no current limit pole of switch  80 . 
     As taught above, when the instantaneous load current is less than the limit current i lim , comparator  83  opens switch  79  and links the output of module  77  to summer  84  thereby affecting normal controller operation where the system drives the system output frequency f eout  toward the commanded frequency f*. Here, summer  84  and integrator module  78  operate to expedite the frequency following process and the output of summer  84  is passed on to limiter  82 . Limiter  82  maintains the output frequency f eout  within an acceptable range (i.e., between f elow  and f ehigh ). 
     Referring yet again to FIG. 5, when the instantaneous load current (i.e., i sf ) is greater than limit current i lim , comparator  83  closes switch  79  to link the output of proportional scalar module  70  to one input of summer  86  and switches the output of scalar module  76  to the input of summer  84 . The output of summer  84  is provided as a second input to summer  86 . The output of summer  86  is provided to limiter  82  and the output of limiter  82  is provided to boost limiter  46  (see again FIG.  2 ). Thus, it should be appreciated that the frequency based current limiter, as its name implies, adjusts the output frequency as a function of the difference between a feedback stator current i sf  and the limit current i lim  in an attempt to maintain the stator current below the limit value i lim . 
     Referring now to FIG. 6, one exemplary embodiment of the torque boost limiter  46  includes first and second square modules  100  and  102 , first through fourth summers  104 ,  108 ,  122  and  126 , one square root module  106 , first and second multiplier  110  and  128 , respectively, a proportional gain module  112 , an absolute value module  114 , a sign module  116 , first and second normally open (NO) contacts  118  and  120  and a limiter module  124 . The current limit value i lim  is squared by module  100  and its output is provided to summer  104 . Similarly, the synchronous d-axis feedback signal i def  is squared by module  102  and its output is provided to summer  104 . Summer  104  subtracts the output of module  102  from the output of module  100  and provides the different to square module  106  which, as its label implies, provides the square root of the received value as an output i qemax . Thus, the output of module  106  corresponds to a maximum synchronous q-axis current value i qemax  given the current limit value i lim  and the synchronous d-axis feedback signal i def  fed to modules  100  and  102 . If maximum value i qemax  is exceeded an excess current condition will likely occur. 
     Absolute value module  114  receives the synchronous q-axis feedback current i qef  and provides the absolute value thereof to summer  108 . Summer  108  subtracts the absolute value of feedback current i qef    114  from the maximum synchronous q-axis current i qemax  and provides a q-axis difference value Δi qe  as an output to multiplier  110 . Q-axis difference value Δi qe , like difference value Δi s  in FIG. 5, indicates an excessive current condition. To this end, where q-axis difference value Δi qe  is positive, the instantaneous q-axis feedback current is less than maximum value i qemax  and no limit condition exists (i.e., a current tripping condition does not exist). However, where q-axis difference value Δi qe  is negative, the instantaneous q-axis feedback current i qef  is greater than maximum value i qemax  and an excessive q-axis current condition exists. 
     Sign module  116 , as its label implies, determine the sign of q-axis feedback current i qef  and provides that sign as an input to multiplier  110 . Multiplier  110  multiplies difference value Δi qe  and the sign of the q-axis feedback current i qef  and provides the result as an input to proportional scalar module  112 . Module  112  multiplies its input by proportional gain factor K p  and provides its output to first contact  118 . Upon examination of the calculations performed by the upper portion of limiter  46  as illustrated in FIG. 6, it should be appreciated that the input to contact  118  will always be a value that tends to drive the q-axis current i qe  toward the maximum q-axis value i qemax . For instance, in the case of positive rotation motoring where q-axis current i qef  exceeds maximum value i qemax , the value provided to contact  118  will be negative. Similarly, during positive motoring where q-axis current i qef  is less than maximum value i qemax , the value provided to contact  118  will be positive. Other scenarios with negative motoring, positive generation and negative generation are contemplated. 
     Contact  118  is controlled by the sign of difference value Δi qe . Where the sign of value Δi qe  is positive (i.e., i qef  is less than i qemax ), contact  118  remains open and the nominal torque boost value determined by module  44  is not altered. However, where q-axis difference value Δi qe  is negative (i.e., i qef  is greater than i qemax ), contact  118  is closed to facilitate reduction of boost value R s i qef  as described below. 
     The output of contact  118  is provided to contact  120 . Contact  120  is controlled as a function of the output frequency f eout  such that the torque boost limiter is only activated when the output frequency f eout  is at a relatively small fraction of a motor nameplate frequency (e.g., 0.06 p.u.). Thus, in the present example, where output frequency f eout  is greater than 0.06 p.u. the nameplate frequency contact  120  remains open and where f eout  is less than 0.06 p.u. of the nameplate frequency contact  120  is closed. 
     Referring still to FIG. 6, summer  122  receives the nominal torque boost R s i qef  from the nominal torque boost module (see FIG. 4) and adds the nominal torque boost R s i qef  to the output of contact  120  (i.e., either a zero value if either of contacts  118  or  120  or both contacts  118  and  120  are open or the stepped up signed value K p Δi qe ) and provides its output to limiter module  124 . Limiter module.  124  limits the voltage boost term to between zero and the nominal torque boost value R s i qef  and provides a limited voltage boost value V lb  as an input to summer  126 . 
     Referring yet again to FIG. 6, multiplier  128  multiplies the rated flux current λ rated  and 2π times the electrical output frequency f eout  thereby generating the reactive voltage drop value 2πf e λ rated . The reactive value is provided as a second input to summer  126 . Summer  126  adds the limited voltage boost value V lb  and the reactive voltage drop thereby generating the synchronous q-axis voltage value V qe  which is provided to transformer  50  as illustrated in FIG.  2 . 
     B. Experimental Results 
     Referring now to FIG. 7, waveforms corresponding to measured current data for two of three motor phases generated without the inventive system are illustrated. It can be seen that, relatively quickly, under the circumstances tested, the w-phase motor current value exceeded a current limit reference value i lim  and the current tripping mechanism of the motor control system is activated thereby cutting off current to the motor. 
     Referring to FIG. 8, waveforms corresponding to two-phase currents that are similar to the waveforms of FIG. 7 are illustrated, albeit generated using a controller employing the inventive torque/voltage boost limiting method. To this end, comparing FIGS. 7 and 8, it can be seen that the current trip in FIG. 7 is avoided and instead, the inventive system used to generate the waveform of FIG. 8 simply and smoothly adjusts both system frequency f e  and the commanded system current to avoid a trip condition. 
     C. Other Embodiments 
     A method similar to that described above can be applied in the case of a simple V/f controller where a system operator or user sets a low speed voltage boost directly via an adjustable user parameter (e.g., via a user interface—see again FIG.  1 ). Here, when the voltage boost parameter is set too high, a current trip condition can occur. 
     Referring to FIG. 9, a voltage-frequency curve is illustrated that shows operation of a typical simple V/f drive. Here, it can be seen that there are generally three separate zones of operation corresponding to three differently sloped sections of characteristic curve. The three zones include a first zone between zero frequency and a break point frequency (e.g., 0.06 p.u. the nameplate frequency), a second zone between the breakpoint frequency and the nameplate frequency and a third zone above the nameplate frequency. 
     As in the case above, in the present case, the invention is provided to kick in at low operating frequencies and hence is only concerned with system operation between zero and the breakpoint frequency. Again, at higher operating frequencies it is assumed that the frequency limiting scheme described above (see again FIG. 5) will limit current and avoid current tripping conditions. 
     In this simplified system type, the voltage reference equations can be expressed as:                V   qe     =       V   boost     +       (         V   BP     -     V   boost         f   BP       )          f   eout                 Eq.  3                 V   de     =   0           Eq.  4                                
     where V boost  is the user set boost voltage value, V BP  is the breakpoint voltage (see again FIG. 9) and f BP  is the breakpoint frequency. 
     Referring now to FIG. 11, a second simplified q-axis voltage reference generator  150  is illustrated. In FIG. 11, the generator includes four summers  162 ,  168 ,  152  and  160 , one divider  164 , one multiplier  166 , one scalar module  154  and two contacts  156  and  158 . Here a user supplies each of the breakpoint voltage and frequency values as well as a desired voltage boost value V boost  and the current limit value i lim . In addition, the output frequency f eout  is obtained from limiter  82  in FIG. 5 and a feedback current i sf  can be obtained from the output of square root module  64  in FIG.  5 . 
     Summer  162  adds the breakpoint voltage V BP  and boost voltage value V boost  and provides the sum to divider module  164 . Divider module  164  divides the sum from summer  162  by the breakpoint frequency f BP . Multiplier  166  multiplies the output of divider  164  by output frequency f eout  to provide the reactive second term in Equation 3 above. Next, summer  168  adds the voltage boost value V boost  to the output of multiplier  166  thereby completing Equation 3 and generating an un-limited q-axis voltage value V qe . 
     Referring still to FIG. 11, summer  152  subtracts the feedback current i sf  from the current limit value i lim  to generate a difference value Δi s  which is stepped up by module  154  and is provided to contact  156 . Contact  156  is similar to contact  118  in FIG. 6 except that contact closure is conditioned upon stator current difference value Δi s  instead of q-axis current difference value Δi qe . Thus, contact  156  closes when value Δi s  is negative and remains open when stator current difference value Δi s  is positive. Contact  158  operates in a fashion that is identical to contact  120  in FIG.  6 . 
     Summer  160  adds the output of summer  168  and contact  158 . The value of the signal from contact  158  is always negative or zero and therefore summer  160  either leaves the V qe  value unchanged or reduces the value at low frequencies f eout  and when the limit current i lim  is exceeded. 
     Although not illustrated in FIG. 11, it is contemplated that the frequency based current limiter of FIG. 5 or some configuration similar thereto would operate along with the FIG. 11 configuration. In addition, at frequencies below the breakpoint frequency, the d-axis voltage reference V de  provided to transformer  50  in FIG. 2 is set to zero. 
     A general method  200  according to the present invention is illustrated in FIG.  12 . In FIG. 12, beginning at block  202 , the controller  14  (i.e., a processor within controller  14  running a pulse sequencing program) determines the system operating frequency f e . At block  204 , if the operating frequency is greater than a threshold value (e.g., 0.06 p.u. the rated or nameplate frequency f rated ), control passes to block  206  where the controller  14  operates to limit the q-axis voltage by simply adjusting frequency f e  when necessary. Where f e  is less than the threshold value control passes to block  108  where controller  14  determines if the feedback current (e.g., i qef  or i sf ) is less than the corresponding current limit (e.g., i lim  in the case of i sf  and i qemax  in the case of i qef ). Where the feedback current is less than the limit current control passes to block  206  and V qe  is limited by controlling frequency f e . However, at block  208 , where the feedback current exceeds the limit or maximum current, control passes to block  210  where controller  14  reduces boost value R s i qe  to maintain the system current below a trip condition. 
     While the invention as described above in the context of an exemplary method and apparatus, it should be appreciated by those skilled in the art that the present invention contemplates other embodiments and therefore should not be limited by the description above and instead, the claims that follow hereafter should be referred to determine the scope of the invention. For example, while the invention is described as one wherein either a q-axis current feedback or a stator feedback current is used to determine when the voltage boost value should be altered, it should be appreciated that current derivatives such as a filtered current feedback signal may be employed instead of a pure q-axis or stator feedback current value. Other modifications to the embodiment above are contemplated. 
     To apprise the public of the scope of this invention, the following claims are made.