Abstract:
Reference signal generators using thermistors are disclosed. An apparatus includes a first device having a first temperature coefficient and a thermistor having a second temperature coefficient having a sign opposite to that of the first temperature coefficient. A circuit maintains equivalence of a first signal and a second signal and offsets a first temperature variation of the first device using a second temperature variation of the thermistor to generate the second signal having a low temperature coefficient. The first device may be a bipolar transistor configured to generate a base-emitter voltage and coupled in series with the thermistor. The first signal may be a first voltage on a first node. The second signal may be a second voltage on a second node. The circuit may be configured to maintain effective equivalence of the first voltage and the second voltage. The apparatus may include a resistor coupled to the second node.

Description:
BACKGROUND 
     1. Field of the Invention 
     The present invention relates to integrated circuits and more particularly generating a reference signal in integrated circuits. 
     2. Description of the Related Art 
     In general, a bandgap reference circuit provides a voltage reference with improved temperature stability and is less dependent on power supply voltage than other known voltage reference circuits. Bandgap reference circuits typically generate a reference voltage approximately equal to the bandgap voltage of silicon extrapolated to zero degrees Kelvin, i.e., V G0 =1.205V. Typical voltage reference circuits include a current minor coupled to the power supply and the voltage reference node to provide a current proportional to absolute temperature (i.e., PTAT) to the voltage reference node. These circuits can be made with relatively low cost, but have the disadvantages of being sensitive to mechanical strain and/or aging, which reduces the accuracy of the voltage reference. In addition, typical voltage reference circuits generate PTAT (or equivalent) output currents that vary across temperature, which make those voltage references less useful as standalone current generators. An additional voltage-to-current generator is typically used to stabilize the output current. 
     Accordingly, improved techniques for generating reference voltages are desired. 
     SUMMARY OF EMBODIMENTS OF THE INVENTION 
     Reference signal generators using thermistors are disclosed. In at least one embodiment of the invention, an apparatus includes a first device having a first temperature coefficient and a thermistor having a second temperature coefficient. The second temperature coefficient has a sign opposite to a sign of the first temperature coefficient. The apparatus includes a circuit configured to maintain equivalence of a first signal and a second signal and further configured to offset a first temperature variation of the first device using a second temperature variation of the thermistor to generate the second signal having a low temperature coefficient. The first device may be a bipolar transistor configured to generate a base-emitter voltage. The thermistor may be coupled in series with the bipolar transistor. The first signal may be a first voltage on a first node, and the second signal may be a second voltage on a second node. The circuit may be configured to maintain effective equivalence of the first voltage and the second voltage. The apparatus may include a resistor coupled to the second node and having a third temperature coefficient. The third temperature coefficient may have a magnitude substantially less than a magnitude of the first temperature coefficient and substantially less than a magnitude of the second temperature coefficient. The first signal may be a first current and the second signal may be a second current. The first device may be a metal-oxide-semiconductor field-effect transistor (MOSFET) device coupled to the thermistor and coupled to a second MOSFET device having a different gate-to-source voltage than the first MOSFET device. 
     In at least one embodiment of the invention, a method includes. maintaining equivalence of a first signal and a second signal. The method includes offsetting a temperature variation of a third signal having a first temperature coefficient using a thermistor having a second temperature coefficient to generate the second signal having a low temperature coefficient. Maintaining equivalence may include generating an indicator of a voltage difference between a first voltage on a first node coupled to a first load including a series combination of the thermistor having a resistivity proportional to temperature and a diode having the first temperature coefficient. The second signal may be a second voltage on a second node coupled to a second load. The method may include adjusting the first and second signals in response to the indicator. The method may include controlling a first current source to generate the first voltage in response to the indicator. The method may include controlling a second current source to generate the second voltage in response to the indicator. The first signal may be a first current and the second signal may be a second current. Offsetting the temperature variation may include using the thermistor to compensate for a difference in gate-to-source voltages of a first metal-oxide-semiconductor field-effect transistor (MOSFET) device and a second MOSFET device. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The present invention may be better understood, and its numerous objects, features, and advantages made apparent to those skilled in the art by referencing the accompanying drawings. 
         FIG. 1  illustrates a voltage reference generator circuit. 
         FIG. 2  illustrates voltage as a function of temperature for various nodes of the voltage reference generator circuit of  FIG. 1 . 
         FIG. 3  illustrates a voltage reference generator circuit having lower thermal noise than the voltage reference generator circuit of  FIG. 1 . 
         FIG. 4  illustrates an integrated circuit polysilicon resistor. 
         FIG. 5  illustrates an integrated circuit diffusion resistor. 
         FIG. 6  illustrates a voltage reference generator circuit including a thermistor consistent with at least one embodiment of the invention. 
         FIG. 7  illustrates an integrated circuit metal resistor. 
         FIGS. 8A and 8B  illustrate an integrated circuit silicided polysilicon resistor. 
         FIG. 9  illustrates a V GS -R voltage reference generator circuit including a thermistor consistent with at least one embodiment of the invention. 
         FIGS. 10A and 10B  illustrate various embodiments of a current reference generator circuit including a thermistor consistent with at least one embodiment of the invention. 
     
    
    
     The use of the same reference symbols in different drawings indicates similar or identical items. 
     DETAILED DESCRIPTION 
     A typical bandgap voltage reference utilizes temperature behavior of diodes to generate a voltage having a negative temperature coefficient (i.e., a negative first-order temperature coefficient) and a voltage having a positive temperature coefficient (i.e., a positive first-order temperature coefficient) and combines those voltages to produce an approximately zero temperature coefficient reference voltage. In general, voltage reference circuits take advantage of two electrical characteristics to achieve the desired V REF : the V BE  of a bipolar transistor is nearly complementary to absolute temperature, e.g., V BE =(−1.5 mV/°K*T+1.22)V, and V T  is proportional to absolute temperature, i.e, V T =kT/q. Although pure diodes are preferable because they generate a higher diode drop for the same current, the typical bandgap voltage reference manufactured in a complementary metal-oxide-semiconductor (CMOS) process uses diode-coupled, bipolar junction transistors (i.e., BJTs or bipolar transistors), which are readily available in a CMOS process (e.g., PNPs for instance are bipolar devices formed from P-diffusion, an N-well, and a P-well in a CMOS process). The voltage across the diodes (or diode-coupled bipolar junction transistors) has a negative temperature coefficient, but the voltage difference between two diode drops in which the current densities differ is proportional to absolute temperature (PTAT). The use of two banks of bipolar junction transistors of different sizes (or two identical banks with different currents) can generate ΔV BE . The typical bandgap forces ΔV BE  across a relatively temperature insensitive resistor (e.g., a polysilicon resistor) using negative feedback, which generates a PTAT current through the resistor. Another resistor is placed in series, which amplifies ΔV BE  to cancel the negative temperature coefficient of the diode drop. 
     Referring to  FIG. 1 , a typical voltage reference circuit (e.g., voltage reference generator  100 ) provides a temperature stable reference voltage, V REF . A voltage proportional to absolute temperature (i.e., a PTAT voltage) may be obtained by taking the difference between two V BE s biased at different current densities: 
                 Δ   ⁢           ⁢     V   BE       =       V   T     ⁢     ln   ⁡     (       J   1       J   2       )           ,         
where J 1  and J 2  are the current densities of corresponding bipolar transistors. Accordingly, voltage reference circuit  100  includes a pair of PNP bipolar transistors (i.e., transistors  106  and  108 ) that are coupled in a diode configuration (i.e., the collectors and bases of these transistors are coupled together) and coupled to ground. Transistor  108  has an area that is N times larger than the area of transistor  106 . Thus, the current densities of transistor  108  and transistor  106  vary by a factor of N. The emitter of transistor  106  is coupled to an inverting input of operational amplifier  116 . The emitter of transistor  108  is coupled, via resistor R 1 , to the non-inverting input of operational amplifier  116 . Operational amplifier  116  maintains equivalent voltages at nodes  118  and  120 , i.e., V 118 =V 120 =V BE106 . Hence, the difference between V BE106  and V BE108  (i.e., ΔV BE106,108 ) forms across resistor R 2 . Operational amplifier  116  and transistors  102  and  104  convert this voltage difference into a current (i.e., current I PTAT ) proportional to the voltage difference:
 
               I   PTAT     =         Δ   ⁢           ⁢     V       BE   ⁢           ⁢   106     ,   108           R   2       =         V   T     ⁢     ln   ⁡     (   N   )           R   2               
Since the thermal voltage V T  is proportional to absolute temperature via the constant factor k/q, k=1.38*10 −23 J/K and q=1.6*10 −19 C, the current proportional to the voltage difference is also proportional to an absolute temperature, i.e., I PTAT  is a PTAT current.
 
     Transistor  108  provides a voltage nearly complementary to absolute temperature (i.e., a ‘CTAT’ voltage) because the V BE  of a bipolar transistor is nearly complementary to absolute temperature. By compensating the PTAT current with a CTAT voltage, transistors  102 ,  104 ,  106 , and  108 , and resistors R 1  and R 2 , may be appropriately sized to generate a particular reference voltage output having an approximatley zero temperature coefficient: 
                 V   REF     =       V     BE   ⁢           ⁢   106       +       MV   T     ⁢     ln   ⁡     (   N   )             ,         where   ⁢           ⁢   M     =       R   1     /     R   2         ;                     V   REF     =       0.74   ⁢           ⁢   V     +         1.5   ⁢           ⁢   mV       °   ⁢           ⁢   K       ⁢   T         ;                 at   ⁢           ⁢   300   ⁢   °   ⁢           ⁢   K     ,       V   REF     =         0.74   ⁢           ⁢   V     +     0.45   ⁢           ⁢   V       =       1.19   ⁢           ⁢   V     ≈     1.2   ⁢           ⁢     V   .                   
V REF  is approximately equal to, V G0 =1.205V, i.e., the bandgap voltage of silicon extrapolated to zero degrees Kelvin.
 
     Adding a PTAT voltage to a diode drop produces an approximately zero temperature coefficient point at approximately 1.2 V, resulting in a circuit that is not substantially sensitive to the effects of process variation on the bipolar junction transistor. The ratiometric manner in which the resistors are used also reduces effects of process variation, aging, and strain sensitivity. However, a noise transfer function of voltage reference circuit  100  is dependent on a ratio of the load resistors. In an exemplary embodiment of the voltage reference, the ratio of R 1  to R 2  is approximately 5 to 10 (i.e., R 1 /R 2 ≈5-10), and ΔV BE , which is typically less than 100 mV, is amplified along with its noise by operational transconductance amplifier  116 . Operational transconductance amplifier  116  has a feedback factor of R 2 /(R 1 +R 2 ), which causes a reduction in loop gain and bandwidth from the open loop gain. 
     A technique for reducing effects of noise on the reference voltage as compared to noise sensitivity of a reference voltage generated by voltage reference generator  100  includes using a V GS -R topology. For example, the voltage reference generator of  FIG. 3  uses the zero temperature coefficient point of device  320 , i.e., the point where a constant current causes no change in the V GS  of device  320  due to cancellation of the negative temperature coefficient of the threshold voltage of device  320  with the overdrive voltage (i.e. V DSAT ) positive temperature coefficient of device  320 . This circuit forces the voltage across a resistor with substantially no temperature coefficient to be equal or approximately equal to the zero temperature coefficient V GS  of device  320 . Although this circuit has lower thermal noise as compared to the circuit of  FIG. 1 , the circuit of  FIG. 3  is sensitive to process variations since the reference voltage can be affected by the threshold voltage, resistance, mobility, oxide capacitance, and transistor dimensions. In general, the threshold voltage of a MOSFET is particularly sensitive to process variations. In addition, the load of circuit  300  has flicker noise that may be difficult or expensive to reduce or eliminate. For example, the flicker noise may be reduced by increasing the area of device  320  or by increasing V REF , which effectively requires increasing the threshold voltage of device  320 . 
     Referring to  FIGS. 3 and 4  conventional CMOS analog circuits and bandgap voltage reference circuits use polysilicon resistors. Polysilicon resistors typically have highly linear resistances and are designed to have small temperature coefficients. However, polysilicon resistors are sensitive to aging and strain due to their polycrystalline structure. Referring to  FIG. 5 , typical CMOS processes also include diffusion resistors, which are less commonly used due to their large voltage and temperature coefficients. Diffusion resistors are also considered piezoresistive, i.e., sensitive to strain. Although the voltage reference generators that use thin film polysilicon resistors or diffusion resistors when building a bandgap or V GS -R reference circuit are relatively low cost, the response of those resistors to mechanical strain and/or aging degrades the accuracy of the reference voltage. In addition, more power efficient references require lower noise alternatives to satisfy specifications of associated application. Those circuits may generate PTAT (or similar) output currents that vary with temperature variation, making them less useful as standalone current generators. Therefore, an additional voltage-to-current generator may be included to stabilize the output currents. 
     Referring to  FIG. 6 , a V BE -R thermistor  voltage reference generator technique generates a reference voltage by forcing a constant current into a series combination of a thermistor  422  having a resistance R thermistor , and a diode, e.g., diode-coupled bipolar junction transistor  420 . The current is determined by comparing the voltage across that load with a voltage across polysilicon resistor, e.g., polysilicon resistor  424 , having a resistance, R constant , that is constant with respect to temperature variation. This topology has a lower operational transconductance amplifier noise transfer function than conventional bandgap voltage reference generators and has a more consistent output voltage than a V GS -R voltage reference generator. In addition, the load of a V BE -R thermistor  voltage reference generator generates little or no flicker noise. Thus the only substantial sources of flicker noise are operational transconductance amplifier  116  and one or more current sources included in voltage reference generator  400  (e.g., devices  102  and  104 ). The operational transconductance amplifier noise can be reduced or eliminated using chopping or other techniques known in the art. The current sources of a V BE -R thermistor  voltage reference generator have larger noise transfer functions than current sources in conventional bandgap voltage reference generators and may dominate the noise. Nonetheless, the V BE -R thermistor  voltage reference generator topology may offer a lower thermal noise floor at the same power consumption as conventional voltage reference generators. In at least one embodiment of a V BE -R thermistor  voltage reference generator, only one current source is used (e.g., the two current sources are combined using a single device). For example, the drain of the current source node is coupled to two paths including resistors and/or cascode devices, or other suitable circuit elements, that allow operational transconductance amplifier  116  to receive a differential signal that may be processed by the feedback loop to establish a valid operating point. As a result, any variation in the current source current does not affect the ratio of currents in the two branches of the load, the stable operating point does not change, so the feedback maintains V REF  at its original value. Thus, the loop gain of the feedback suppresses noise and the noise would be zero in an infinite gain system. 
     By making use of thermistors, i.e., resistors that have a resistance that varies substantially with temperature, e.g., PTAT metal resistors and/or PTAT silicided resistors, in the core of the voltage reference generator, only a constant current may be generated and provided to R thermistor  to maintain a zero temperature coefficient on V REF . Accordingly, a voltage-to-current generator that generates a constant current that may be required by other voltage reference generator topologies can be eliminated when there is no need for alterative circuits elsewhere in the system. In addition, since metal and silicide resistors are not piezo-resistive, the associated voltage reference generator response has little or no strain sensitivity. Moreover, aging of these types of metal and silicide resistors is generally superior to alternative integrated circuit resistors, increasing stability of the output voltage as a function of time. Another benefit of embodiments of the V BE -R thermistor  voltage reference generator includes lower noise than conventional voltage reference generator topologies. 
     Metal resistors are not commonly used in conventional analog circuits and bandgap voltage reference circuits since metal layers in typical CMOS processes are intended to provide low-resistance interconnects and thus have very low sheet resistance. The low sheet resistance (e.g., 60 milli-Ohms per square) requires resistors having large area to implement even small resistances (e.g., 10-20 kilo-Ohms) However, a stack of multiple metal layers coupled by conductive via(s) of a CMOS process may be configured as electrically coupled metal resistors (e.g.,  FIG. 7 ) that have reduced area as compared to a typical CMOS resistor, (e.g., a planar resistor formed using a narrow, serpentine metal trace implemented using a single CMOS metal layer). In at least one embodiment, the thermistor comprises a metal resistor having a resistance, R thermistor , of approximately zero Ohms 
     Referring to  FIGS. 6, 8A, and 8B , in at least one embodiment, thermistor  422  includes silicided-polysilicon resistors, which are polysilicon resistors without the silicide blocked. Silicide is metal that is injected into the top of polysilicon or diffusion to decrease the sheet resistance. This means that thermistors of silicided-polysilicon resistors have a combination of polysilicon and metal resistor properties, which makes them close to a PTAT resistor. Silicided-polysilicon resistors are less sensitive to strain and aging than conventional CMOS resistors. Typical silicided-polysilicon resistors have higher sheet resistances than metal resistors (e.g., 10 times the typical sheet resistance of metal) and result in metal resistors with higher resistances for the same area (e.g., 100-200 kilo-Ohms) 
     Referring back to  FIG. 6 , although thermistor  422  is illustrated as a single metal resistor, in other embodiments of a V BE -R thermistor  voltage reference generator, thermistor  422  includes a network of individual thermistor elements and/or includes one or more silicided-polysilicon resistors. In at least one embodiment, the circuit of  FIG. 6  may be used as a temperature sensor or as a combination voltage reference generator and temperature sensor by providing a temperature-varying signal from the load (e.g., V TVAR ). Note that in other embodiments, V TVAR  may be the voltage drop across R met  or a combination of the V BE  of diode-coupled bipolar junction transistor  420  and the voltage drop across R met . Other embodiments of a V BE -R thermistor  voltage reference generator combine the metal resistor with a polysilicon resistor to form one composite resistor with an arbitrary first-order temperature coefficient. The composite resistor embodiment of a V BE -R thermistor  voltage reference generator allows generation of a constant reference voltage at a voltage other than the bandgap voltage of silicon. In addition, the composite resistor embodiment of a V BE -R thermistor  voltage reference generator may be exploited for generation of an arbitrary first-order temperature coefficient current. 
     Referring to  FIG. 9 , in at least one embodiment, a reference generator having a V GS -R topology includes a thermistor and is configured to generate a current, I, that is constant with respect to temperature variations, i.e., has current with an approximately zero temperature coefficient. By including a thermistor with a positive temperature coefficient, the voltages across the resistor and device  320  can be higher than in typical V GS -R reference generators and circuit  900  has improved thermal noise as compared to a current generator using circuit  300  of  FIG. 3 . 
     Referring to  FIGS. 10A and 10B , reference generator  1000  generates a current that has a low or approximately zero temperature coefficient. Devices  1002  and  1004  have different sizes, but are biased with the same gate voltage. Since devices  102  and  104  are matched, the currents through devices  102  and  104  are equal to I BIAS . Reference generator  1000  achieves a stable operating point when the voltage drop across the thermistor compensates for the difference in the gate-to-source voltages of devices  1002  and  1004 . This topology is may be used with a resistor having no temperature coefficient where the difference in the gate-to-source voltages of devices  1002  and  1004  are used to generate a bias signal with a constant transconductance. By using a thermistor, the circuit may be used as a constant current reference without a bipolar junction transistor. 
     Circuits  1000  are less sensitive to process variations than the V GS -R topology described above since the threshold voltage does not affect the bias current. In addition, circuit  1000  can operate at a lower supply voltage since 1.2V is not required to produce a bandgap voltage. Circuit  1000  is simpler than other reference generator circuits since the circuit behaves as an amplifier and an operational transconductance amplifier is not required. However the output currents of circuits  1000  may include flicker noise and may be noisier than the output of a V GS -R reference, but not as noisy as a bandgap voltage reference generator. Circuits  1000  are strain insensitive. Note that circuits  1000  do not generate a voltage with a zero temperature coefficient since the constant current that flows through the thermistor results in a temperature dependent voltage. The temperature coefficient of the thermistor should be less than a metal resistors PTAT resistivity. Accordingly, the thermistor may be implemented using a polysilicon resistor in series with a metal resistor to obtain the target temperature coefficient. Note that circuits  1000  and the other self-biased circuits described herein require a startup circuit to prevent the circuit from latching in an off state. Any suitable startup circuit known in the art may be used. 
     The description of the invention set forth herein is illustrative, and is not intended to limit the scope of the invention as set forth in the following claims. For example, while the invention has been described in an embodiment in which p-type MOSFETs are configured as current sources and a PNP-type bipolar junction transistor is used to generate the V BE , one of skill in the art will appreciate that the teachings herein can be utilized with n-type MOSFETs configured as current sinks and an NPN-type bipolar junction transistor coupled to generate the V BE . In addition, diodes may be stacked to further enhance ΔV BE  (e.g., for embodiments including two diodes stacked in series for each bipolar device, ΔV BE  becomes V T In(N 2 )). Variations and modifications of the embodiments disclosed herein, may be made based on the description set forth herein, without departing from the scope and spirit of the invention as set forth in the following claims.