Abstract:
LO leakage and Image are common and undesirable effects in typical transmitters. Typically, thirty complex hardware and algorithms are used to calibrate and reduce these two impairments. A single transistor that draws essentially no de current and occupies a very small area, is used to detect the LO leakage and Image Rejection signals. The single transistor operating as a square law device, is used to mix the signals at the input and output ports of the power amplifier (PA). The mixed signal generated by the single transistor enables the simultaneous calibration of the LO leakage and Image Rejection.

Description:
CROSS-REFERENCE TO OTHER APPLICATIONS 
       [0001]    Not Applicable. 
       BACKGROUND OF THE INVENTION 
       [0002]    Federal Communications Commission (FCC) has allotted a spectrum of bandwidth in the 60 GHz frequency range (57 to 64 GHz). The Wireless Gigabit Alliance (WiGig) is targeting the standardization of this frequency band that will support data transmission rates up to 7 Gbps. Integrated circuits, formed in semiconductor die, offer high frequency operation in this millimeter wavelength range of frequencies. Some of these integrated circuits utilize Complementary Metal Oxide Semiconductor (CMOS), Silicon-Germanium (SiGe) or GaAs (Gallium Arsenide) technology to form the dice in these designs. This standard is called the IEEE 802.11ad protocol. 
         [0003]    The transmit path of the signal being, transferred in the wireless channel in these communication system need to be compensated for various mismatch conditions occurring in the up-convertor circuit. Some of these conditions manifests as LO leakage and signal image in the transmitter RF signal spectrum. 
         [0004]    CMOS (Complementary Metal Oxide Semiconductor) is the primary technology used to construct, integrated circuits. N-channel transistors and P-channel transistors (MOS transistor) are used in this technology which uses fine line technology to consistently reduce the channel length of the MOS transistors. Current channel lengths are 40 nm, the power supply of VDD equals 1.2V and the number of layers of metal levels can be 8 or more.  
         [0005]    CMOS offers the computing power to perform many of the required compensation techniques to overcome the adverse conditions in the transceiver. Yet, the computing power must be used in a power efficient manner to insure that the dissipated power is low enough to allow these important building blocks of the transceiver fabricated in CMOS to be used in mobile applications. This helps to insure that the energy drawn from the limited power contained in the battery is minimized while achieving the optimum performance. 
       BRIEF SUMMARY OF THE INVENTION 
       [0006]    Various embodiments and aspects of the inventions will be described with reference to details discussed below, and the accompanying drawings will illustrate the various embodiments. The following description and drawings are illustrative of the invention and are not to be construed as limiting the invention. Numerous specific details are described to provide a thorough understanding of various embodiments of the present invention. However, in certain instances, well-known or conventional details are not described in order to provide a concise discussion of embodiments of the present inventions. 
         [0007]    One embodiment relates to the use of a single transistor to perform the mixing of two RF spectra in a series signal path and down-convert the signal to extract signal components that indicate the level of LO leakage and image rejection within the spectra of the desired RF signal. Previous techniques to perform the same function required complex circuit components, such as a mixer, a feedback VCO, and a filter in order to detect the distortions. These three major processing blocks require dozens of transistors, inductors, capacitors and can occupy a significant portion of area on the integrated circuit substrate (chip). The VCO alone requires a charge pump, a loop filter, a pre-scalar, a divider, a crystal oscillator and/or a sigma delta modulator. Furthermore, these circuit components need to be designed for stability concerns where the PLL transient behavior, settling time, VCO capacitor bank calibration etc. within the feedback VCO needs to be designed and simulated to operate within tight tolerances. A single transistor replaces all of these components simplifying the design, reducing the area and power dissipation by nearly two orders of magnitude. 
         [0008]    Another embodiment relates to the tapping of the RF spectra from different pairs of ports in a series signal path. The single transistor performs the mixing function can be coupled to any two ports of the series signal path to determine the down-converted signal components. Furthermore, the two tapped ports can be flipped and applied to the single transistor where the transistor can still operate to detect the level of LO leakage and image rejection within the spectra of the desired RF signal. 
         [0009]    Another embodiment relates to an apparatus to mix a first signal with a second signal comprising: a plurality of circuit elements coupled in series forming a series signal path; one of the plurality of circuit elements having an input node and an output node coupled within the series signal path; a gate of a transistor connected to the input node; a source of the transistor connected to the output node; and a drain of the transistor coupled to a resultant node, wherein the transistor mixes the first signal at the input node with the second signal at the output node a generating a mixed signal between the first signal and the second signal at the resultant node, further comprising: a first spectra comprising a first homodyne signal, a first LU leakage signal, and a first image rejection signal applied to the input node; and a version of the first spectra comprising to second homodyne signal, a second LO leakage signal, and a second image rejection signal modified by the one of the circuit elements at the output node, further comprising: an input port coupled to an input of the series signal path; and an output port coupled to an output of the series signal path, wherein an up-converted RF signal is coupled to the input port, further comprising: a low pass filter (LPF) coupled to the resultant node; and a digital signal processor (DSP) coupled to the LPF, wherein the DSP calculates correction factors to reduce the LO leakage signals and the image rejection signals in all the spectra. The apparatus further comprising: an antecedent circuit element coupled to the one of the circuit elements at the input port; and a subsequent circuit element coupled to the one of the circuit elements at the output port, wherein an up-converted RF signal is coupled to the antecedent circuit element, wherein the one of the circuit elements is an amplifier stage that either non-inverts or inverts the second signal with regard to the first signal, wherein the one of the circuit elements is an amplifier stage that amplifies and phase shifts the second signal with regard to the first signal.  
         [0010]    Another embodiment relates to an apparatus to generate a self-mixed signal comprising: a first circuit element including an input node and an output node; a gate of a transistor coupled to the input node; a source of the transistor coupled to the output node; a drain of the transistor coupled to a resultant node; a first spectra comprising a first homodyne signal, a first LO leakage signal, and a first image rejection signal applied to the input node; and a version of the first spectra comprising a second homodyne signal, a second LO leakage signal, and a second image rejection signal modified by the circuit element and generated at the output node, wherein the transistor mixes the first spectra with the version of the first spectra generating the self-mixed signal at the resultant node, further comprising: an antecedent circuit element with an input port coupled to first circuit element at the input node; and a subsequent circuit element coupled with an output port coupled to the first circuit element at the output node, wherein an up-converted RF signal is coupled to the input port, further comprising; an antenna coupled to the output port. The apparatus further comprising: a low pass filter coupled to the resultant node, wherein the first circuit element is an amplifier stage that either non-inverts or inverts the input signal at the output node, wherein the first circuit element is an amplifier stage that amplifies and phase shifts the input signal at the output node. 
         [0011]    Another embodiment relates to a method to generate a mixed signal between two selected ports comprising the steps of coupling a plurality of circuit elements in series forming a series signal path; assigning separate ports between two adjacent circuit elements within the series signal path, wherein an input port couples to an input of a first circuit element in the series signal path and an output port couples to an output of a last circuit element in the series signal path; selecting any two of the ports; connecting a gate of a transistor to a first selected port of the two of the ports; connecting a source of the transistor to a second selected port of the two of the ports; and coupling a drain of the transistor to a resultant node, wherein the transistor mixes a signal at the first selected port with a signal at the second selected port, thereby generating at the resultant node the mixed signal between the two selected ports, wherein the signal at the first selected port has a first spectra comprising at least one of a homodyne signal, a LO leakage signal, and an image rejection signal applied to the input node; and the signal at the second selected, port has a second spectra comprising at least one of a version of the homodyne signal, a version of the LO leakage signal, and a version of the image rejection signal modified by at least one of the plurality of circuit elements, wherein the version components of the second spectra comprises at least a non-inverted or an inverted, an amplified or an attenuated, or a phase shifted component of the first spectra, further comprising the steps of coupling a low pass filter to the resultant node, further comprising the steps of: coupling a digital signal processor (DSP) to an output of the low pass filter to calculate correction factors to reduce the LO leakage signals and the image rejection signals of all signals within the series signal path. The method further comprising the steps of: coupling an up-convened RF signal is to the input port, wherein the first selected port corresponds to an input node or an output node of an amplifier, and the second selected port corresponds to a remaining node of the amplifier. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0012]    Please note that the drawings shown in this specification may not necessarily be drawn to scale and the relative dimensions of various elements in the diagrams are depicted schematically. The inventions presented here may be embodied in may different forms and should not be construed as limited to the embodiments set forth herein. Rather, these embodiments are provided so that this disclosure will be through and complete, and will fully convey the scope of the invention to those skilled in the art. In other instances, well-known structures and functions have not been shown or described in detail to avoid unnecessarily obscuring the description of the embodiment of the invention. Like numbers refer to like elements in the diagrams, unless noted otherwise. 
           [0013]      FIG. 1A  depicts a baseband processor—transmitter—antenna path. 
           [0014]      FIG. 1B  illustrates a direct conversion transmitter driven by I/Q signals 
           [0015]      FIG. 2A  shows a direct. conversion up-converter driven by I/Q signals along with phase/amplitude/DC offset distortions and corrective circuitry in accordance with an embodiment of one of the present invention. 
           [0016]      FIG. 2B  depicts the spectra the circuit of  FIG. 2A  generates in accordance with an embodiment of one of the present invention. 
           [0017]      FIG. 2C  shows a direct conversion up-converter driven by a transmitter digital to analog converter (DAC) I/Q signals along with phase/amplitude/DC offset distortions and corrective circuitry in accordance with an embodiment of one of the present invention. 
           [0018]      FIG. 3A  shows a direct conversion transmitter driven by I/Q signals along with compensation adjustments for phase and amplitude distortions and a detection circuitry based around the power amplifier (PA) in accordance with an embodiment of one of the present invention. 
           [0019]      FIG. 3B  illustrates the down-converted spectra in the circuit of  FIG. 3A  assuming the input spectra of  FIG. 2A  in accordance with an embodiment of one of the present invention. 
           [0020]      FIG. 4  presents a circuit schematic of the power amplifier in accordance with an embodiment of one of the present invention. 
           [0021]      FIG. 5  depicts in a block diagram of the detection circuit connected to the power amplifier to reduce distortions in accordance with an embodiment of one of the present invention. 
           [0022]      FIG. 6A  shows a single transistor operating as the detecting element in the detector circuit in accordance with an embodiment of one of the present invention. 
           [0023]      FIG. 6B  illustrates another connectivity for a single transistor operating as the detecting element in the detector in accordance with the present invention. 
           [0024]      FIG. 7  presents another set of tap points coupling the transmitter path to the detecting circuit in accordance with the present invention.  
       
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       [0025]    The WiGig standard transmits signals in the 60 GHz range of carrier frequencies allowing nearly a 10 GHz signal bandwidth capability. This is a direct conversion system which is known to suffer from I/Q mismatch causing images to form in the spectra of the transmitted signal. This is due to the gain and phase distortion mismatch between the I (real) and the Q (90 phase shifted) signal paths. A desirable feature is to perform a calibration procedure to eliminate distortions caused by the gain and phase distortions. 
         [0026]    Another distortion occurs when the oscillator signal leaks into the signal path before being unconverted. The oscillator signal when mixed with itself causes a DC offset to occur in the signal path. Additional DC offset occurs due to component mismatches within the mixer itself. This additional DC offset causes local oscillator (LO) leakage when the mixer mixes the LO signal. This DC offset can saturate the following stages and needs to be reduced. A desirable feature is to perform another calibration procedure to eliminate the distortions caused by this DC offset. 
         [0027]    A baseband-transmitter block diagram is illustrated in  FIG. 1 . The signal enters the baseband processor  1 - 1  and is processed in preparation to be transmitted into free space via an antenna. The processing can occur according to one of many given standards known in the industry. The processed signal is converted to analog with as digital to analog converter D/A  1 - 2 . The analog signal from the D/A is applied to the analog transmitter which tip-converts the baseband signal to a higher carrier frequency. One example is a direct conversion transmitter where the spectrum of the baseband signal is translated by the local oscillator to the RF carrier frequency using a quadrature converter. The unconverted signal is transferred to the antenna  1 - 4  and sent into free space. 
         [0028]    Further details of the analog transmitter  1 - 3  are illustrated in  FIG. 1B . The inputs are quadrature baseband signals: Q sig  and I sig , where each the I sig  has a phase difference of  90  from that of Q sig . These orthogonal signals, if combined, contain the original information. These two signals each also comprise the identical spectra. The Q sig  is applied to the mixer  1 - 7 , while the IL sig  is applied to the second mixer  1 - 6 . Both mixers are switched by the quadrature outputs of the local oscillator (LO)  1 - 5 . The cos ω osc  t waveform is used to mix Q sig , while the sin ω osc  t waveform is used to mix I sig . The outputs of the mixers  1 - 6  and  1 - 7  are added together by adder  1 - 8 . The combined signal is applied to the pre-amplifier  1 - 9  which then drives the power amplifier (PA)  1 - 10 . The PA  1 - 10  further amplifies the signal. A matching network  1 - 11  insures that the maximum power from the PA is transferred to the antenna  1 - 4 . The up-converted signal is provided with sufficient power for propagation into free space. 
         [0029]      FIG. 1B  illustrates a transmitter with ideal I/Q signals (phase and amplitude constant over the bandwidth) and an ideal oscillator which does not leak signal from the LO  1 - 5  into the baseband signal paths. 
         [0030]    However,  FIG. 2A  illustrates the transmitter with various forms of distortion introduced into the I/Q signals such as phase/amplitude distortion and an effective DC offset caused by LO carrier leakage and circuit mismatch. For example, the Q sig  input applied to the input of the mixer  1 - 7  contains the desired signal Q sign  plus the undesired phase/amplitude distortions and a DC offset; similarly, the I sig  input applied to the input of the mixer  1 - 6  contains the desired signal I sig  plus the undesired phase/amplitude distortions and a DC offset These undesired distortions can degrade the ideal transmitted signal characteristics sought after in the idea circuit of  FIG. 1B . 
         [0031]    The distortions in the amplitude and phase of the I/Q signals occur, in part, because of the frequency dependence of the transfer functions used to generate the I/Q signals, leakage of signals in parasitic, and transistor and component mismatch. For example, the DC offset occurs, in part, because mixers  1 - 7  and  1 - 6  mix the LO signals with leaked carrier LO signals in the signal path. This causes the transmitter output to contain a portion of the unmodulated LO carrier and an image signal. 
         [0032]    The up-converted spectra  2 - 1  at the output of the adder  1 - 8  is presented in  FIG. 2B . The spectrum of the desired single side band signal is ω sig    2 - 4 . The spectrum of the undesirable LO leakage signal  2 - 3  is shown at ω lo . The phase and gain mismatch in the I/Q signals causes the spectrum of the undesired image tone of the signal  2 - 2  at ω img . The undesired image tone is offset below the LO leakage signal as the desired single sideband signal is above the LO leakage signal. The tones are separated by δF which is the frequency delta between (ω lo -ω img ) and (ω sig -ω lo ). In a typical transmitter, these 3 tones (ω img , ω lo , ω sig ) are propagated, through a series signal path of the Pre-amp  1 - 9  PA  1 - 10 , Matching Network  1 - 11 , Antenna  1 - 4  chain and become components of the RF output. It is highly desirable to calibrate the transmitter so that the tones ω img  and ω lo  are minimized or, if possible, eliminated. 
         [0033]    The carrier leakage signal ω lo    2 - 3  typically occurs in the analog baseband segment of the transmitter. The quadrature signal suffers carrier leakage due to acquired DC offsets in the signal path that combines within the transmitter signal path causing the signal to contain the unmodulated carrier. The unmodulated carrier is the source of the carrier leakage signal and generates a distortion in the desired signal, since the carrier leakage is transmitted with the desired signal. As the power of the desired signal is reduced due to system requirements, the carrier leakage signal may dominate the overall signal. Therefore, it is very desirable to reduce the carrier leakage to improve the quality of the desired signal. 
         [0034]    The I/Q mismatch signal in direct conversion systems can degrade the signal quality of the desired signal ω sig    2 - 4 . The mismatch occurs within the quadrature paths of the baseband segment of the transmitter. The I/Q components of the quadrature signal each carry a given bandwidth of signal information. Ideally, it is desirable if the characteristics of the circuitry in the baseband segment of the I/Q signal paths exactly match each other over the entire bandwidth of signal information. In this ideal situation, the I/Q mismatch would be reduced to near zero values, since there would be a 90° phase shift and equivalent magnitudes between the corresponding components of the I and Q signals. 
         [0035]    However, the actual characteristics of the circuitry in the baseband segment typically do not match each other over the entire bandwidth of the spectra carrying the signal information. Typically, the phase and amplitude of the I/Q signals are matched at the center of the bandwidth of signal information. Since the signal information has a bandwidth centered on the carrier frequency,  some of the signal information components are located away from the center carrier frequency. Since the components forming the I/Q paths are not matched at these frequencies away from the carrier frequency, the I/Q signals carried within these frequencies are typically processed with different phase and amplitude characteristics. Thus, the way the I signal is processed at an frequency offset of δω from the carrier frequency by the circuitry may not match the way the Q signal is processed at an frequency offset of δω from the carrier frequency by the circuitry. This is known as I/Q mismatch and occurs between the I/Q paths within the bandwidth of signal information. The result of this mismatch causes an unwanted sideband image ω img    2 - 2  to be generated with the signal spectra as illustrated in  FIG. 2B . The I/Q mismatch introduces distortion into the spectra of the desired signal and causes the constellation of the modulated signal being transmitted to be distorted. Therefore, it is very desirable to reduce the I/Q mismatch to improve the quality of the desired signal. 
         [0036]    Given that the amplitudes of the offending spectra of carrier leakage and the I/Q mismatch signals need to be reduced; the first step is to detect these offending spectra due to the mismatches. Once the offending spectra is detected, various circuit techniques and algorithms can be used together to reduce the offending spectra and thereby improve the signal quality of the desired signal. 
         [0037]    The algorithm in conjunction with various circuit configurations can be implemented in a computer. The algorithm may also contain instructions that, when executed, perform one or more methods, such as those described above. The information carrier is a computer- or machine-readable medium, such as the memory, the storage device, or memory on processor. 
         [0038]    These computer programs (also known as programs, software, software applications or code) include machine instructions for a programmable processor, and can be implemented in a high-level procedural and/or object-oriented programming language, and/or in assembly/machine language. As used herein, the terms “machine-readable medium” “computer-readable medium” refers to any computer program product, apparatus and/or device (e.g., magnetic discs, optical disks, memory, Programmable Logic Devices (PLDs)) used to provide machine instructions and/or data to  a programmable processor, including a machine-readable medium that receives machine instructions as a machine-readable signal. The term “machine-readable signal” refers to any signal used to provide machine instructions and/or data to a programmable processor. 
         [0039]    One example of a detection circuit  2 - 12  is illustrated in  FIG. 2A  and is used to detect the offending spectra. This version of the detection circuit requires at least three major processing blocks: a mixer  2 - 5 , a feedback VCO  2 - 6 , and a filter  2 - 7  in order to detect the distortions. These three major processing blocks require dozens of transistors, inductors, capacitors and can occupy a significant portion of area on the integrated circuit substrate (chip). The VCO alone requires a charge pump, a loop filter, a pre-scalar, a divider, a crystal oscillator and/or a sigma delta modulator. Furthermore, the detection circuit  2 - 12  as illustrated in  FIG. 2A  needs to be designed for stability concerns where the PLL transient behavior, settling time, VCO capacitor bank calibration etc. within the feedback VCO  2 - 6  needs to be designed and simulated to operate within tight tolerances. 
         [0040]    In addition to the area usage, these processing blocks dissipate power. For a portable system, a battery can provide a given amount of energy between recharges. These processing blocks drain the energy from the battery and require the battery to be charged between uses at shorter time intervals. 
         [0041]    The output of the adder  1 - 8  is applied to the mixer  2 - 5  in the detection circuit  2 - 12  and mixed with the signal from a feedback VCO  2 - 6 . The output of the mixer  2 - 5  is filtered by the filter  2 - 7  and applied to the input of the ADC  2 - 8 . Once the detected signal is filtered  2 - 7 , the filtered signal is converted into the digital domain by the ADC and processed by an algorithm programmed with the digital signal processor (DSP)  2 - 9 . Once these distortions are detected, the measured values are used to decrease the amount of distortion by using a feedback circuit to minimize each of the distortion components. 
         [0042]    For example, a number of calibration techniques can be used such as the least mean square (LMS) algorithm. Several measurement tests are typically performed by the DSP  2 - 9  to adjust the amplitude, the phase, and the DC offset such that the distortion due to I/Q mismatch and carrier leakage are each minimized. The DSP uses various algorithms based on these calibration techniques to measure and adjust these parameters. The algorithms may be programmed using software programs, computer code, machine code, etc. 
         [0043]    As each of these calibrations are performed, an adjustment block  2 - 10  adjust the adjustable components (not shown) within the LO  1 - 5  signal path and in the adjustable components (not shown) within the I/Q signal paths. These adjustments reduce the undesired LO leakage and image rejection distortions. The measurements can be performed iteratively during inactive periods and the results can be stored in memory. Then, the frequency of the feedback VCO  2 - 6  can be altered to mimic a different carrier frequency and perform the detection, measurements, and adjustments again such that the distortions at this different carrier frequency can be minimized. 
         [0044]    Note that the tap point of the measurement occurs after the adder  1 - 8 . Thus, this feedback correction does not compensate for any additional distortion that may occur in the series signal path containing the pre-amp  1 - 9  and PA  1 - 10  illustrated in  FIG. 1B . 
         [0045]      FIG. 2C  illustrates a correction technique where the correction signal from adjustment block  2 - 10  is applied to the adders  2 - 16  and  2 - 17 . In this case, the correction signals are added to the digital stream of the I signal  2 - 19  and the Q signal  2 - 18  in the digital domain. The I/Q signal are then applied to the digital to analog converters  2 - 15  and  2 - 14 , respectively. The analog I/Q signals now contain the compensated phase/amplitude and DC offset to offset the undesired phase/amplitude and DC offset signals. The feedback loop containing the up-converter, the detection circuit, ADC, and DSP monitor the detected signal and can iteratively apply correction signals to the adders  2 - 16  and  2 - 17  until the distortion has been minimized or eliminated. 
         [0046]      FIG. 3A  illustrates an innovative embodiment of detecting the offending spectra without the use of a feedback VCO  2 - 6  or mixer  2 - 5  as illustrated in  FIG. 2A . Instead, the detection circuit  3 - 1  taps two ports from the series signal path of a series network within the dotted block  3 - 7 . One preferred embodiment is the tapping of the ports  3 - 3  and  3 - 4 , as depicted in  FIG. 3A . A series signal path is defined as a signal path that comprises one or more plurality of circuit elements couple in series. Separate ports are assigned between two adjacent circuit elements within said series signal path. The input to the series signal path is labelled as the input port, while the output of the series signal path is labelled as the output port. The circuit elements can comprise: pre-amp, amplifiers, low noise amplifier, certain filters, matching networks, etc. One example is the path illustrated comprising the nodes  3 - 2 ,  3 - 3 ,  3 - 4 , and  3 - 5  of such a path; however, communication system circuits can comprise many such series signal paths within an integrated circuit chip. For example. receive path in a transceiver containing a low noise amplifier, pre-amp and amplifier would be another example of a series signal path. 
         [0047]    One particular series signal path comprises the path formed by coupling the pre-amp  1 - 9 , the PA  1 - 10 , and the matching network  1 - 11  serially coupled and presented within the dotted block  3 - 7 . The series signal path is tapped at two ports and applied to the detection circuit  3 - 1 . In this case, at the port  3 - 3  applying the RF signal RF  in  to the PA and port  3 - 4  of the PA generating the RF signal RF out  at the output of the power amplifier (PA)  1 - 10 , although other tapped ports from the series signal path could be used. The spectra of these RF signals at each of the ports in this series path are similar to that illustrated in  FIG. 2B . At least two different ports of the RF spectra from the series path are applied to the detection circuit  3 - 1 . 
         [0048]    These two ports of  3 - 3  and  3 - 4  of the PA  1 - 10  circuit element are applied to the detection circuit  3 - 1  which contains a squaring function capability. Note that the signal at port  3 - 4  is an amplified version of the signal at port  3 - 3 ; thus, the spectra at port  3 - 3  will also be amplified at the port  3 - 4 . The spectra at port  3 - 4  are a version of the spectra at port  3 - 3 . The squaring circuit multiplies the two versions of the spectra of the RF signal by itself. The signals from these tapped ports are effectively mixed against each other, thereby eliminating the need for the feedback VCO  2 - 6  and the mixer  2 - 5 . The components of the spectra at each of the tapped ports are given in EQU. 1 (disregarding the amplification). 
         [0000]        I   ω   =A   1  cos(ω 1   t );  L   ω   =A   2  cos(ω 2   t ); and  S   ω   =A   3  cos(ω 3   t )  (EQU. 1)
 
         [0049]    The detection circuit  3 - 1  in  FIG. 3A  generates the spectra illustrated in  FIG. 3B  at resultant node  3 - 6 . 
         [0050]    The output of the squaring transistor is given in EQU. 2: 
         [0000]      ( I   ω   +L   ω   +S   ω )*( I   ω   +L   ω   +S   ω )=[ A   1  cos(ω 1   t )+ A   2  cos(ω 2   t )+ A   3  cos(ω 3   t )] 2   (EQU. 2)
 
         [0000]    and expanded in EQU. 3: 
         [0000]      =( A 1)( A 2) cos)ω 1   t −ω 2   t )+( A 1)( A 3) cos(ω 1   t−ω   3   t )+( A 2)( A 3) cos(ω 2   t−ω   3   t )+ . . . other terms  (EQU.3)
 
         [0000]    In EQU. 3, the “other terms” contain the DC term and higher order frequency terms of ((ω 1 t+(ω 2 t)), ((ω 2 t+(ω 3 t)), etc. A low pass filter  2 - 11  is used to filter out these higher order frequency terms. The filtered squaring function output spectra  3 - 7  at the output of the LPF  2 - 11  are depicted in  FIG. 3B , where T 1    2 - 15  is the tone consisting of [(A1)(A2) cos(ω 1 t−ω 2 t)+(A2)(A3) cos(ω 2 t−ω 3 t)] and T 2    2 - 16  is the tone consisting of (A1)(A3) cos(ω 1 t−ω 3 t). The DC term is presented as DC  2 - 14 . 
         [0051]    Note that the tone T 1    2 - 15  is located at a frequency separated from DC  2 - 14  by δF  2 - 17  and the tone T 2    2 - 16  is located at a frequency separated from DC  2 - 14  by 2*δF  2 - 18 , This is due to the squaring function within the detection circuit  3 - 1  that multiplies the spectra illustrated in  FIG. 2B  by another version of the spectra modified by at least one circuit element. The calibration algorithm then adjusts the DC offset of the transmitter, and the gain/phase of the transmitter I/Q path to reduce these two tones: T 1  and T 2 . When T 1  and T 2  are reduced to minimum, the undesirable RF output of the LO leakage and image tones are also reduced to a minimum. 
         [0052]      FIG. 4  illustrates one embodiment of the circuit schematic  4 - 6  of the series signal path  3 - 7  for the transmitter stage. The pre-amp  1 - 9  and the PA  1 - 10  in the dotted box  3 - 7  of  FIG. 3A  is illustrated as transistors, capacitors, and inductors. Transistors N 1  and N 2  are sized to provide a scaled amplification. The transistor nomenclature for the N 1  and N 2  transistors imply a N-channel MOS (Metal Oxide Semiconductor) transistor. The pre-amp comprises the inductor L 1    4 - 3  and the transistor N 1    4 - 1 , while PA  1 - 10  comprises the inductor L 2    4 - 4  and the transistor N 2    4 - 5 . The Pre-amp is coupled to the PA via the capacitor  4 - 2  The signal from the adder  1 - 8  is coupled to the gate of transistor N 1  at port  3 - 2 . The pre-amp amplifies the signal and couples the amplified signal to the gate of transistor N 2  at port  3 - 3 . The PA signal at port  3 - 4  is coupled to the matching network  1 - 11 . The matching circuit can be designed to transfer maximum power between the port  3 - 4  and port  3 - 5  coupled to the antenna  1 - 4 . The spectra as illustrated in  FIG. 2B  can be found at the ports  3 - 2 ,  3 - 3 ,  3 - 4  and  3 - 5  in varying amplitude levels (depending, on the gain of the pre-amp, PA, and matching network). 
         [0053]      FIG. 5  depicts the ports  3 - 3  and  3 - 4  of the series signal path being tapped and coupled to the detection circuit block  3 - 1  which multiplies the spectra at port  3 - 3  with the spectra at port  3 - 4 . Since these spectra are similar but vary in amplitude, the multiplier is called a squaring function. The result of the squaring function is coupled to the output of the detection circuit at resultant node  3 - 6 . The spectra of the detected signal at resultant node  3 - 6  is processed as before by the LPF, ADC, DSP, etc. to adjust the phase, amplitude and DC offset so that T 1  and T 2  are minimized. 
         [0054]      FIG. 6A  presents one embodiment of the inventive use of a single transistor  6 - 1  which can be used to perform the squaring function. Transistor N 3  can be a very small device, of the order of 1/1100 the size of the power amplifier (PA) transistor N 2 . The source (S) of transistor N 3  is coupled to port  3 - 3  of the series path while the gate (G) of N 3 is coupled to port  3 - 4  of the series path. The substrate (SUB) of transistor N 3  can be tied to ground, although other voltage potential levels can be used. Due to the device being small. N 3  has negligible impact on normal operation of the PA circuit since the loading of the transistor N 3  on the port  3 - 3  and  3 - 4  of transistor N 2  are minimal. In addition, the transistor N 3  behaves as a square law device and multiplies the spectra at port  3 - 3  by the spectra at port  3 - 4  and generates the output at the drain (D) of transistor N 3 which is the resultant node  3 - 6 . Thus, the single transistor N 3  performs the function performed by the mixer  2 - 5  and the feedback VCO  2 - 6  in the detection circuit  2 - 12  of  FIG. 2A . The spectra of the detected signal at drain node (D) of transistor N 3  is processed as before by the LPF, ADC, DSP, etc. to adjust the phase, amplitude and DC offset so that T 1  and T 2  are minimized. 
         [0055]    Furthermore, since drain of N 3  is connected to gates of transistors in the LPF  2 - 11 , there is no de bias current through the N 3  device. N 3  operates like a passive mixer, where both the gate and source nodes of the transistor are connected to the RF output signal ports in the series signal path carrying the spectra of the RF signals: I ω =A 1  cos(ω 1 t), L ω =A 2  cos(ω 2 t), and S ω =A 3  cos(ω 3 t). As a passive mixer, it multiplies the RF spectra times a version of itself producing the down-converted signal illustrated in  FIG. 3B  and described by the earlier equation EQU. 3. 
         [0056]    This embodiment of detection circuit  3 - 1  can save over two orders of magnitude in chip area real estate and power dissipation when compared to the complex circuit components and design of the mixer  2 - 5  and feedback VCO  2 - 6  presented in the detection circuit  2 - 12  of FEC.  2 C. In addition, the issues of stability and other design concerns for the VCO and associated circuitry in the detection circuit  2 - 12  are eliminated, simplifying the overall design of the inventive detection circuit which utilizes a single transistor. 
         [0057]      FIG. 6B  presents another embodiment of the inventive use of a single transistor  6 - 2  which can be used to perform the squaring function. Transistor N 4  can be sized the same as the transistor  6 - 1  in  FIG. 6A . However, the source (S) of transistor N 4  is coupled to port  3 - 4  of the series path while the gate (G) of N 4  is coupled to port  3 - 3  of the series path. The substrate (SUB) of transistor N 4  can be tied to ground, although other voltage potential levels can be used. Similarly, N 4  has negligible impact on normal operation of the PA circuit since the loading of the transistor N 4  on the ports  3 - 3  and  3 - 4  of transistor N 2  are minimal. In addition, the transistor N 4  behaves as a square law device and multiplies the spectra at port  3 - 3  by the spectra at port  3 - 4  and generates the output at the drain (D) of transistor N 4  at resultant node  3 - 6 . Thus, the single transistor N 4  performs the function performed by the mixer  2 - 5  and the feedback VCO  2 - 6  in the detection circuit  2 - 12  of  FIG. 2A . The spectra of the detected signal at drain node (D) of transistor N 4  is processed as before by the LPF, ADC, DSP, etc. to adjust the phase, amplitude and DC offset so that T 1  and T 2  are minimized.  
         [0058]      FIG. 7  depicts a circuit similar to that of  FIG. 5  except the series signal path is tapped at ports  3 - 2  and  3 - 3  and applied to the detection circuit  3 - 1 . Other possible tapped port pairs in the series signal path can include:  3 - 2 ,  3 - 4 ;  3 - 3 ,  3 - 5 ; etc. Since the tapped ports are applied to gate and drain of the transistor in the detection circuit  3 - 1 , a voltage different between the tapped points generates a (V g -V s ) voltage which can be used to generate an I ds  current. The detection circuit can utilized the transistor configuration as illustrated in either  FIG. 6A  or  FIG. 6B . The spectra of the detected signal at drain node (D) of transistor within the detection circuit  3 - 1  is processed as before by the LPF, ADC, DSP, etc. to adjust the phase, amplitude and DC offset so that T 1  and T 2  are minimized. 
         [0059]    This embodiment is innovation since the transistors N 1  or N 4  in the detection circuit  3 - 1  eliminates the need to be concerned with the details of the VCO, the charge pump, loop filter, pre-scalar, divider, crystal oscillator and/or sigma delta modulator design or specifications of these components. Nor is there a need to describe the PLL transient behavior, settling time, VCO capacitor hank calibration etc. 
         [0060]    Finally, it is understood that the above descriptions are only illustrative of the principle of the current invention. Various alterations, improvements, and modifications will occur and are intended to be suggested hereby, and are within the spirit and scope of the invention. This invention may, however, be embodied in many different forms and should not be construed as limited to the embodiments set forth herein. For example, the various embodiments presented can be used for any of the various wired or wireless standards incorporating a series signal path within a transceiver. These techniques can be employed on the receive or transmit paths to extract information from a series signal path. Rather, these embodiments are provided so that the disclosure will be thorough and complete, and will fully convey the scope of the invention to those skilled in the arts. It is understood that the various embodiments of the invention, although different, are not mutually exclusive. In accordance with these principles, those skilled in the art may devise numerous modifications without departing from the spirit and scope of the invention. Although N-MOS transistors were used in the circuit schematics, P-MOS transistors can be easily be designed to perform similar capabilities. In addition, a network and a portable system can exchange information wirelessly by using communication techniques such as Time Division Multiple Access (TDMA), Frequency Division Multiple Access (FDMA), Code Division Multiple Access (CDMA), Orthogonal Frequency Division Multiplexing (OFDM), Ultra Wide Band (UWB), WiFi, WiGig, Bluetooth, etc. The network can comprise the phone network, IP (Internet protocol) network, Local Area Network (LAN), ad hoc networks, local routers and even other portable systems.