Abstract:
A multi-stage integrator achieves a relatively high small-signal gain, broad bandwidth, and very clean transient pulse response. Only simple inverters are used, making the design scalable to deep sub-micron with low supply voltages, a rail-to-rail output swing, and a relatively low output impedance and useful tolerance to capacitive loading. A high gain amplifier is coupled between an integrator input node and amplifier output node. A broadband transconductor is coupled between the integrator input node and integrator output node. A resistor connects the amplifier output node and the integrator output, while a capacitor is coupled from the integrator input to the amplifier output. The conductance of the resistor (the reciprocal of the resistance, or 1/R) is selected to be substantially equal to the transconductance g m  of the transconductor. A method for achieving clean transient pulse response is also described.

Description:
FIELD OF THE INVENTION 
     This invention relates generally to operational amplifiers and in particular to CMOS operational amplifiers, and is more particularly directed toward a multi-stage integrator having high gain and fast transient response. 
     BACKGROUND OF THE INVENTION 
     An op amp (operational amplifier) architecture is desirable which is suited to current and foreseeable future generations of small geometry CMOS (complementary metal-oxide-semiconductor), manufactured economically in high volume using the same processes as those used for manufacturing digital circuitry. 
     The conventional op amp, illustrated in FIG. 1 in block diagram form, and generally depicted by the numeral  100 , comprises two gain stages. The first functions as a differential transconductance (g m ) stage  101  and the second as an integrator  103 , separated by a differential to single-ended converter  102 . The conventional op amp  100  is illustrated in more detail in FIG.  2 . 
     As shown in FIG. 2, the g m  stage  101  comprises a differential pair  201 ,  202  with a single current source “tail”  203  (both typically, and as an example, p-type insulated-gate field effect transistors), and two current source loads  204 ,  205  (typically, and as an example, provided by n-type transistors). By selecting an output  206  from only one of the differential input stages, differential to single-ended conversion is accomplished or, conventionally, current sources  204  and  205  are implemented as a mirror with single-ended output  206  derived from the high impedance side of the mirror. 
     This single-ended output  206  is then applied to the integrator stage  103 . In the implementation shown, the integrator  103  includes a n-type output transistor  207  with a current source tail  210 , and Miller capacitor  208 . A nulling resistor  209  has been added for the sake of stability. 
     In sub-micron CMOS technology, it is difficult to achieve an integrator with a combination of high gain and wide bandwidth with a high slew rate and a good transient response to high frequency events. The active devices are fast, but a single gain stage has very low DC gain. This may be increased by techniques such as cascading, but to a limited extent; also, deep sub-micron processes have very restricted supply voltages which make it desirable to use the full voltage range efficiently without cascoding. A multi-stage integrator (typically three inverting gains) gives high gain with simple inverters, but must be stabilized with an internal nested pole, which sharply degrades the bandwidth and thus results in a poor slew rate and poor transient response. 
     Consequently, a need arises for an integrator implementation that provides high gain and good transient response, while offering simplicity of design and economy in overall circuit area. 
     SUMMARY OF THE INVENTION 
     These needs and others are satisfied by the present invention, in which a three-stage integrator achieves a high small-signal gain on the order of 80 dB, with 200 MHz typical bandwidth, and very clean transient pulse response. Only simple inverters are used, making the design scalable to deep sub-micron with low supply voltages, a rail-to-rail output swing, and a relatively low output impedance and useful tolerance to capacitive loading. 
     In accordance with one aspect of the invention, a high-gain, fast response amplifier comprises a first amplifier path including a plurality of inverters and a first amplifier path output, a second amplifier path having a common input with the first amplifier path, and including a transconductor and a second amplifier path output, and a resistor interconnecting the first and second amplifier paths to form a composite amplifier having the common input as the input thereto and the output of the second amplifier path as the output thereof. 
     The first amplifier path may include first and second cascaded inverters coupled to an output stage, while the second cascaded inverter may include a compensation network connected in feedback to improve stability. The compensation network may be a series RC network. In a preferred form of the invention, the amplifier further comprises a capacitor connected in feedback around the first amplifier path. The resistor interconnecting the first and second amplifier paths preferably has a conductance g equal to the transconductance g m  of the transconductor. 
     In accordance with a further aspect of the present invention, a method is provided for minimizing high-frequency transient signals at an amplifier output. The method comprises the steps of providing an amplifier having an amplifier input and amplifier output, feeding back a sample of high frequency output signals from the amplifier output to the amplifier input, providing a secondary signal path having a common input with the amplifier input, and adding the secondary signal path output to the amplifier output to effectively remove the high-frequency transient signals. 
     In accordance with yet another aspect of the present invention, an integrator comprises a first signal path including multiple cascaded inverting amplifiers coupled between the integrator input and an amplifier output, a second signal path limited to a single transconductor coupled between integrator input and output, a resistor coupled between the amplifier output and the integrator output, and a capacitor coupled between the amplifier output and the integrator input. The first signal path provides a relatively high-gain, narrowband amplifier, and the second signal path provides a relatively low-gain, broadband amplifier, and the first and second signal paths sum through the resistor to form a single amplifying structure with relatively high low-frequency gain, and relatively fast high-frequency transient response. 
    
    
     Further objects, features, and advantages of the present invention will become apparent from the following description and drawings. 
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 depicts a conventional operational amplifier of the prior art in block diagram form; 
     FIG. 2 is a more detailed schematic representation of the op amp of FIG. 1; 
     FIG. 3 depicts an integrator in accordance with the present invention; 
     FIG. 4 illustrates the integrator of FIG. 3 in greater detail; 
     FIG. 5 is a device-level schematic diagram of the integrator of FIG. 3; 
     FIG. 6 is a simplified block diagram illustrating a refinement of the integrator of FIG. 3; 
     FIG. 7 is a device-level schematic diagram of the integrator of FIG. 6; 
     FIG. 8 shows gain versus frequency performance of an amplifier suitable for use in an integrator in accordance with the present invention; and 
     FIG. 9 illustrates pulse response timing diagrams for the integrator of FIG.  5 . 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     In accordance with the present invention, a high-gain, fast response integrator is described that offers distinct advantages when compared with the prior art. FIG. 3 depicts an integrator in accordance with the present invention. A high gain amplifier  301  is coupled between an integrator input node  305  and amplifier output node A  306 . A broadband transconductor is coupled between the integrator input node  305  and integrator output node  307 . A resistor  304  connects the amplifier output node A  306  and the integrator output  307 , while a capacitor  303  is coupled from the integrator input  305  to amplifier output  306 . The conductance of the resistor  304  (the reciprocal of the resistance, or 1/R) is selected to be substantially equal to the transconductance g m  of the transconductor  302 . 
     The diagram of FIG. 4 depicts the integrator of FIG. 3 in more detail. As can be appreciated from an examination of FIG. 4, the high gain amplifier  301  comprises simple inverter stages  401  and  402 , which are analog amplifiers also commonly called inverting amplifiers, with a compensation network connected in feedback around inverter  402  to enhance stability. It will be apparent that amplifier  301 , with 3 gain stages in a closed loop formed by capacitor  303 , is inherently unstable. The nested compensation components  404  and  405  make this loop stable according to well-known principles. In the circuit of FIG. 4, the compensation components  404 ,  405  form a series-connected RC (resistor-capacitor) network. 
     The output of inverter  402  drives amplifier output stage  403 , a p-channel enhancement mode MOSFET. The transconductor  302  is an NMOS transistor coupled between the inverter input  305  and the inverter output  307 . As discussed above, resistor  304  is coupled between the high gain amplifier output  306  and the integrator output  307 , with the capacitor  303  coupled between the amplifier output  306  and the integrator input  305  (the capacitor is connected as a feedback element from the amplifier output  306  to the amplifier input  305 ). The value of the resistor  304  is substantially equal to the reciprocal of the transconductance of the transconductor  302 , or 1/g m . 
     Operation of the integrator of FIG. 4 may be understood in one of two ways. First of all, due to the propagation delay through inverter  402 , there will be severe transient ringing at the amplifier output node A  306 . That high frequency voltage will be fed back by capacitor  303  to the input  305 . The input  305  is assumed to be high impedance because, in application, the prior stage is typically a current source output. The voltage at the amplifier output node A  306  will induce a current in resistor  304  given by its conductance g (1/R). The same voltage at the input  305  will induce a current in the transconductor  302  given by its transconductance g m . If g (1/R)=g m , then the integrator output  307  remains unaffected (independent of load capacitance). 
     In the alternative, the circuit may be regarded as two amplifying paths in parallel with common input  305  and output  307 . Amplifier  301 , comprising simple inverters  401  and  402  and output transistor  403 , is a high-gain amplifier with low bandwidth and poor transient response. The transconductor  302  is a low-gain, high bandwidth amplifier with good transient response. The two signal paths, one through the amplifier  301  and the other through the transconductor  302 , sum benignly via resistor  304  to form a single amplifying structure, or composite amplifier, with high low frequency gain and good high frequency transient response. It is noteworthy that the integrator output  307  has a low dynamic output impedance related to the transconductance of transconductor  302  (1/g m ), so the integrator output  307  is relatively tolerant of load capacitance. 
     A device level schematic for implementation of an integrator in accordance with the present invention is shown in FIG.  5 . As noted above, the amplifier  301  comprises three sequential inverters  501 ,  503 ,  403 , made stable by an internal nested compensation resistor  404  and capacitor  405 . The third inverter  403  has (optionally) a class A/B construction to boost the output drive capability. This amplifier has a high gain, greater than 80 dB, typically (small signal), as illustrated by the gain versus frequency performance plot of FIG.  8 . However, due to its nested pole, it has low bandwidth and poor transient response when the loop is closed. The transconductor  302  is a simple NMOS inverting device that has a low voltage gain but very wide bandwidth. 
     The bias voltage applied to transistors  502  and  504  configures them to act as constant current loads to gain devices  501  and  503 . Transconductors  508 ,  509  and  510  are interconnected to form a constant voltage on the gate of device  505 , which biases device  505  such that a proportion of the AC current flowing to the gate of device  403  is diverted to modulate the gate of device  507 , thus establishing a bi-directional push-pull amplifying action. 
     The amplifier  301  (devices  501 ,  503 , and  403 ) has a poor transient response, so that the ultimate effect at node A  306  of an applied input pulse is severe ringing. This ringing voltage causes a current to flow in the resistor  304  (to the integrator output Aout  307 ) proportional to the conductance of the resistor  304 . The conductance of the resistor  304  is the reciprocal of its resistance, or 1/R. 
     This voltage will also be fed back via capacitor  303  to the integrator input  305 , which conventionally will be driven from a high impedance current source, as mentioned previously. The ringing voltage thus appears on the input of the transconductor  302 , and causes a current to flow from the output Aout  307  equal to the transconductance of the transconductor  302 . If g r =g m  the currents cancel and the voltage on Aout  307  is substantially undisturbed by the ringing of the main amplifier. 
     This is illustrated by the pulse response timing diagrams of FIG. 9, that illustrate output waveforms in response to a 5 ns (nanosecond) input pulse. Ringing at the amplifier output is depicted in waveform  901 . The transconductor  302  exhibits a fast transient response, however, so its normal pulse response is as shown in waveform  902 . With feedback as described above, the pronounced ringing response of the main amplifier is effectively cancelled, and the integrator output pulse appears as shown in waveform  903 . 
     A further refinement that may be employed in a practical design is a charge-pumped virtual battery  601  inserted between the transconductor  302  and the resistor  304 , as shown in the simplified diagram of FIG.  6 . When the integrator is used in an application with a unipolar power supply, it is often desired to pull a capacitive load (an analog-to-digital converter, or ADC, for example) to ground, or even slightly below, to achieve zero code output. The simple charge-pumped virtual battery  601  achieves this by maintaining a positive voltage on the transconductor  302  drain with zero output voltage. If the charge pump&#39;s internal resistance rises with increasing output voltage, it does not over-pump the drain voltage and thus achieves an output swing from true zero to near V DD  without increasing the voltage stress on any component beyond V DD —important in a deep sub-micron process. In such a process, a pumping frequency in the GHz (gigahertz) range is practical, which permits pumping and smoothing capacitors of a few pf, which can be integrated on chip. 
     The charge pump  601  advantageously solves the following problem. It may be desirable for device  302  to be able to sink current from the output Aout  307  even when Aout is at ground voltage or even very slightly below. This may be done by means of a power supply or integrated charge pump connected to the source of device  302  to pull it to a significantly negative voltage. 
     However, this technique may cause difficulties in practice. Perhaps the most significant problem is that V SS  and the IC silicon substrate are commonly connected together, which precludes pulling the source of device  302  below ground potential. Secondly, the voltage between V DD  and V SS  may already be at the maximum potential permitted by electrical stress reliability concerns. 
     The operation of the charge pump  601  may be better appreciated through an examination of FIG.  7 . Two clock phases are used, which are conventional non-overlapping 2-phase clocks, labeled  709  (CLOCK) and  710  (CLOCKB). These clocks are used to switch n-type MOS switches  701 ,  702 ,  703 ,  705 . To switch p-type MOS switches  704  and  706  simultaneously with  703  and  705 , a further clock  711  (labeled CLOCKB_P) is used, which has the same timing as CLOCKB but is inverted in polarity. 
     During the first clock phase (CLOCK), capacitor  707  is charged to a fixed voltage that is substantially equal to the supply voltage (V DD —V SS ) less the threshold voltage drop across switch  701 . During the second clock phase (CLOCKB), the capacitor  707  is connected across reservoir capacitor  708 . This produces a charge pumping action such that, as reservoir capacitor  708  is discharged by the current flowing into device  302 , it is replenished from the pump. 
     This circuit has the advantageous feature that device  302  can now sink current from the output Aout  307 , even when Aout is at (or even very slightly below) ground potential, without requiring a negative voltage on the source of device  302 . At the same time, no voltage difference within the circuit is created that exceeds V DD- V SS . 
     There has been described herein a high-gain, fast response integrator that is improved over the prior art. It will be apparent to those skilled in the art that modifications may be made without departing from the spirit and scope of the invention. Accordingly, it is not intended that the invention be limited except as may be necessary in view of the appended claims.