Abstract:
A Transmission Line Pulse (“TLP”) measurement system for testing devices such as integrated circuits (“ICs”), and especially for testing the electrostatic discharge (“ESD”) protection structures connected to terminals on such ICs. The TLP measurement system measures the pulsed voltage and/or current of a device under test (“DUT”) by recording voltage and/or current pulse waveforms traveling in a constant impedance cable to and from the DUT. The pulses going to and returning from the DUT are processed to create signal replicas of the voltage and current pulses that actually occurred at the DUT. Oscilloscope operating settings optimize the recording of these signal replicas by improving the measurement signal-to-noise ratio. This improved TLP system is especially useful when very short width pulses on the order of less than 10 nanoseconds are used to test the DUT&#39;s response.

Description:
CROSS REFERENCE TO RELATED APPLICATION 
   This application claims the benefit of U.S. Provisional Application No. 60/758,248, filed Jan. 11, 2006, which is incorporated herein by reference. 

   BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   This invention generally relates to systems that test integrated circuits and the like, and specifically relates to systems for measuring an integrated circuit&#39;s response to pulses generated by transmission line pulsers. 
   2. Description of the Related Art 
   It is useful to measure the ways integrated circuits (“ICs”) respond to short high power pulses when evaluating their electrostatic discharge (“ESD”) protection. The Transmission Line Pulse (“TLP”) techniques described in the ESD Association Standard Practice ANSI/ESD SP5.5.1-2004, as well as in U.S. Pat. Nos. 5,519,327 and 6,429,674, are often used to make such measurements. Specifically, these techniques are designed to measure the ways ICs respond to current and voltage pulses that are delivered to them by a pulse generator or pulser. The TLP pulser delivers an initial, or incident, TLP pulse through a constant impedance cable to a selected terminal of an IC or other device under test (“DUT”). When the incident TLP pulse reaches the DUT, it is partly reflected by the DUT and new current and voltage pulse waveforms result. The reflected pulse overlaps with the incident pulse as it travels back up the constant impedance cable in the opposite direction, toward the pulser. The relative amplitudes of these incident and reflected pulses are determined by the dynamic impedance of the DUT. The study of a DUT&#39;s response to short high power pulses is a common goal of ESD studies. The constant impedance cable is designed in a manner known in the art to avoid significant pulse distortions, so that the reflected pulse may be accurately measured and the dynamic impedance of the DUT may be calculated by comparing the ratio of the incident and reflected pulses. 
   Prior art circuits for measuring the incident and reflected current and/or voltage pulse waveforms typically use current and/or voltage oscilloscope probes. These probes are positioned in the constant impedance cable at a selected insertion point where the incident and reflected pulses are expected to overlap, which means that the probes need to be positioned only a small distance from the DUT terminal. This approach is particularly useful because when the probes are able to measure the voltage and/or current levels of incident and reflected pulses at a point where the waveforms overlap, it enables measurements to be made that approximate the actual current and/or voltage levels that appear where the cable connects to the DUT terminal, which is the desired measurement point to fully determine the DUT&#39;s response. Additionally, there are also signal measurement techniques well known in the art that improve the accuracy of such measurements but they can only be applied when the pulses overlap. 
   It is known in the art that the actual pulse at the DUT terminal under test is the sum of the incident and reflected pulses, and prior art circuits and devices are not designed to take an actual measurement of these waveforms. This is because prior art circuits measure the waveforms&#39; overlap when the waveforms are slightly displaced in time relative to each other. The waveforms are displaced at the measurement point because prior art designs place their oscilloscope probes a small distance from the DUT, as mentioned above, while the incident and reflected pulses only perfectly overlap at the exact point where the constant impedance cable connects to the DUT. 
   In other words, there will always be a time delay between the incident and reflected pulses when there is any length of cable between the oscilloscope probe and the DUT, and this is a characteristic of prior art devices. The time skew can be very short, but for some TLP measurements a delay as short as one nanosecond will cause measurement errors. This is especially true for cases where the total pulse length is on the order of a few nanoseconds, the skew becomes a significant part of the waveform. 
   DUT response to a TLP pulse can be divided into two time regions, the initial transient response and the steady state response after the transient has dissipated. Prior art TLP systems measure only the steady state response because the overlapped waveforms do not show the overlap of the transient response. When measuring the transient response of the DUT, the first nanosecond is often the most important waveform measurement. The actual DUT waveform thus cannot be recorded using the way in which prior art systems place oscilloscope probes in the constant impedance cable. 
   TLP waveforms are typically recorded with single-shot high-speed digital oscilloscopes. Computers then use the digitized waveform information collected by these oscilloscopes to determine steady state pulse time regions and to calculate the current and/or voltage levels at the DUT by averaging the data in those regions. Currently available oscilloscopes have dynamic ranges limited by their 8-bit analog-to-digital converters to 256 voltage levels and have typical noise levels of four or more least significant bits (limiting signal-to-noise ratios to &lt;64:1). Data averaging improves this limited signal-to-noise ratio, but the resulting signal-to-noise ratio may be inadequate for some applications. Techniques known in the art have been developed to optimize oscilloscope measurements under computer control whereby the oscilloscope input amplifier gain is increased and oscilloscope offset adjustments made to shift the waveform, to zoom in and record the desired measurement region of the waveform. 
   With overlapped incident and reflected pulses, known prior art techniques have controlled oscilloscope gain and offset to improve the signal-to-noise ratio by ten fold. However, when an oscilloscope&#39;s digitization of DUT waveform measurements do not have an overlapped area, or when such overlapped area is small compared to the pulse width, these techniques may not be effective to improve the signal-to-noise ratio. Importantly, when TLP pulse widths are less than 10 nanoseconds (commonly termed Very Fast TLP or “VF-TLP”), very little, if any, overlap of incident and reflected pulses is possible at the measurement point in the constant impedance cable. Without overlap, the incident and reflected pulses are separately recorded and a computer calculates a mathematically generated estimate of the DUT&#39;s current and/or voltage waveforms. 
   The ESD Association Standard Practice ANSI/ESD SP5.5.1-2004 document describes several configurations of TLP. The most commonly used is the Time Domain Refection (“TDR”) configuration. Most TLP systems produce 100 nanosecond-wide pulses. These systems employ oscilloscope measurement probes that usually capture the pulse signals where the incident and reflected pulses overlap in the constant impedance cable. This may be called “TDR-O,” which stands for TDR with overlapped pulses. In contrast, there are TLP systems that measure “TDR-S,” which is TDR using separated pulses where the constant impedance cable is long enough to hold the entire pulse length between the measurement point and the DUT. An advantage of measuring TDR-O is that the oscilloscope control system can optimize the vertical gain, adapting to the signal level of the best use of the oscilloscope&#39;s high-speed digitizer&#39;s dynamic range. 
   As previously mentioned, TLP using pulse widths of 10 nanoseconds or less is often called Very Fast TLP. Due to physical constraints of VF-TLP systems, the current and voltage measurement probes can not be placed close enough to the DUT terminal in the constant impedance cable to allow significant overlap of the incident and reflected pulses as required to make a useful TDR-O measurement. Therefore, TDR-S is the most commonly used configuration for VF-TLP. Unfortunately, oscilloscope gain adaptive gain control is very limited with TDR-S compared with TDR-O. What is needed is a circuit that converts TDR-S signals to TDR-O type signals and thereby resolve this measurement weakness. 
   A drawback of prior art techniques for measuring TDR-O signals is that, although the incident pulse and the reflected pulse overlap, but they are never fully overlapped. As the measurement probes cannot be placed exactly at the point where the transmission cable connects to the DUT terminal, there is always a finite time required for the incident pulse to travel from the measurement point to the DUT terminal. This same amount of time is required for a reflected pulse to travel from the DUT terminal to the measurement probe location. During this down and back travel time, the measurement probe records the incident pulse without the reflected pulse. Then the overlap period is measured, and finally the period where the reflected pulse does not overlap the incident pulse is recorded. Because the overlap is imperfect and the pulses are not perfectly rectangular, or flat topped, the actual DUT terminal current and/or voltage waveforms cannot be directly measured from the recorded signals. Information about the time-varying dynamic transient response of the DUT cannot be clearly established, either. A purpose of the present invention is to obtain fully overlapped pulses in order to provide recording of the undistorted DUT waveforms. These recordings provide both transient and steady state DUT response information. 
   Therefore, a need exits in the art of TLP systems that test the ESD protection of ICs to improve their measurement capabilities by producing a true replica of the DUT electrical signals as they exist at the DUT terminal under test, thereby enabling more accurate measurements of DUT responses to TLP pulses, including measurements of transient responses. 
   SUMMARY OF THE INVENTION 
   Accordingly, it is an object of the present invention to provide improved TLP systems for testing ICs and other devices, especially where there are ESD protection structures in such electronic parts. This invention is a TLP system that measures the current and/or voltage at a DUT terminal due to a pulse delivered through a constant impedance cable or a combination of cable and printed wiring board and measured at points within this constant impedance conduction path in a manner that compensates for the pulse delay caused by the travel time of the pulse from the measurement points to the DUT and back. In addition the new TLP system may include a high-speed oscilloscope and computer with gain optimizing software to improve the measurement&#39;s signal-to-noise ratio. 
   A component of the present invention is a circuit that splits the TDR-S signals into two signal paths. The longer path has a propagation delay that is equal to the shorter path plus twice the measurement probe to DUT propagation time. Signals from the two paths are then combined into a single signal. This combined signal contains separated replicas of three pulses: the incident pulse, followed by the sum of the incident and reflected pulses, further followed by the reflected pulse. The measurement of the current and/or voltage of the DUT can be made on the second pulse in the combined waveform as it is a replica of the DUT electrical waveforms. 
   Broadly stated the present invention is a pulse measurement circuit for measuring the response of an integrated circuit device or the like (a “DUT”) to a pulse generated by a pulse generator wherein said pulse has a predetermined substantially constant voltage and pulse width, comprising: a conductor for coupling said pulse to a selected terminal of said DUT, said conductor having an approximately constant electrical impedance and an approximately constant pulse propagation velocity; a sensing probe connected to said conductor for generating an electrical signal in direct proportion to the current or voltage flowing in said conductor, said sensing probe positioned at an insertion point on said conductor where the propagation time from said insertion point to said DUT terminal is greater than the pulse width of said pulse multiplied by the propagation velocity of said conductor; a signal splitter for dividing the electrical signal from the said sensing probe into two approximately equal sub-signals and for coupling each said sub-signal to a separate output terminal; first and second signal delay paths each having an input end and an output end, each connected at its said input end to a respective one of said signal splitter output terminals, for delaying the propagation time of each said sub-signal a different predetermined amount; a signal combiner coupled to the output end of said first and second signal delay paths for combining said sub-signals into a single output signal; and a recorder for recording said output. 
   An object of the present invention is therefore to provide a signal to a recorder such as a high-speed oscilloscope that is a replica of the pulse generated by the TLP pulser as it exists at the DUT terminal under test. 
   These and other objects, features and advantages of the present invention will no doubt become apparent to those skilled in the art after reading the following detailed description of the preferred embodiments that are illustrated in the several accompanying drawings. 

   
     BRIEF DESCRIPTIONS OF THE DRAWINGS 
     The present invention can be better understood with reference to the attached drawings, which are incorporated in and form a part of this specification, and, together with the description, serve to explain the principles of the present invention. The components within the drawings are not necessarily to scale relative to each other, emphasis instead being placed upon clearly illustrating the principles of the present invention. 
       FIG. 1  is a schematic functional diagram of a transmission line pulse system for testing integrated circuits, including taking measurements of current and voltage signals that are processed by two adder circuits; 
       FIG. 2  is a time diagram of the signals produced by the TLP system that tested the circuits in  FIG. 1 ; and 
       FIG. 3  is a detailed schematic of the preferred embodiment of the invention. 
   

   DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
   The following description provides numerous specific details, such as the identification of various system components, and is designed to offer a thorough understanding of embodiments of the invention. One skilled in the art will recognize, however, that the invention can be practiced without one or more of the specific details, or with other methods, components, materials, etc. In still other instances, well-known structures, materials, or operations are not shown or described in detail to avoid obscuring aspects of various embodiments of the invention. Reference throughout this specification to “one embodiment” or “an embodiment” means that a particular feature, structure, or characteristic described in connection with the embodiment is included in at least one embodiment of the present invention. Thus, the appearance of the phrases “in one embodiment” or “in an embodiment” in various places throughout this specification are not necessarily all referring to the same embodiment. Furthermore, the particular features, structures, or characteristics may be combined in any suitable manner in one or more embodiments. 
   As shown in  FIG. 1 , a pulse is generated by a TLP pulser  10  and transmitted to an input terminal  38  of DUT  40  through a constant impedance path  20 . Path  20  can comprise a conventional cable or other structure for coupling a pulse between two circuits or devices. A current and/or voltage sensing probe  30  is inserted into path  20  to produce signal(s) related to the pulse traveling in path  20 . Where the DUT is a packaged IC, the IC is typically tested in an IC socket and sensing probe  30  is connected to one of the pins of the IC socket. A wafer probe can be used where the DUT is an unpackaged IC. The produced signal(s) detected by the sensing probe  30  are coupled to a pulse adder circuit  50  via a cable  62 . The adder circuit according to the present invention is composed of 4 parts: a signal splitter  52 , two delay cables  54  and  56 , and a signal combiner  58 . The lengths of  54  and  56  are made to have their propagation time difference be extremely close to twice the propagation time between the measurement point where probe  30  is inserted into path  20  and the terminal  38  of DUT  40 . This produces a signal containing a sequence of pulses at the output of the combiner  58 , the second pulse of which is a replica of the pulse that existed at terminal  38 , the point where path  20  connects to DUT  40 . This signal is transmitted by a cable  60  to a recorder  70 , that is commonly a digital oscilloscope (“scope”). Recorder  70  can measure the replicated pulse, thereby determining the electrical conditions that were present at the terminal  38  of DUT  40 . It is useful, but not necessary, to measure both current and voltage in such a manner. Due to the constant impedance of path  20 , called Zo, the current and voltage at the connection of path  20  and terminal  38  of DUT  40  are related by Ohms law: V DUR =I DUT ·Zo. However, the measurement accuracy of the electrical parameters of the DUT is improved by measuring both current and voltage. Therefore, sensing probe  30  is preferably composed of voltage probe  31  and current probe  32  in this embodiment. In the preferred embodiment, a second sensing probe  32 , and a second adder circuit  51 , comprised of splitter  53 , two delay cables  55 ,  57  and a signal combiner  59 , function in an analogous manner as the components of adder circuit  50 , with cables  56  and  57  having lengths as specified below. A cable  63  couples current sensing probe  32  to adder circuit  51  and a cable  61  transmits the signal from the output of  51  to a second input of  70 . The reason a second adder circuit  51  is needed if both current and voltage signals are being measured at the same time is because the voltage and current sensing probes can not physically be inserted on path  20  at the same point because of their size. Consequently, the signal being sensed by each probe needs to be independently tuned by respective adder circuits  50  and  51  since the transit time of pulses between each probe and the terminal  38  of the DUT will be different. 
   More specifically, signals from both cables  60  and  61  are recorded by recorder  70  simultaneously. Having two measurements implies that there must be two different propagation times from probe  31  to DUT  40  and from probe  32  to DUT  40 , as indicated in  FIG. 1 . Adjusting the cable lengths to provide proper delay times means that, with adder circuit  50  processing the voltage waveforms from voltage probe  31  and adder circuit  51  processing the current waveforms from current sensing probe  32 , the delay times Dx through cables  54 ,  56 ,  55  and  57  can be calculated as follows:
 
 D   1   ≈D   3 &gt;2·PulseWidth
 
 D   2   =D   1 +2·( V Transit)
 
 D   4   =D   3 +2·( I Transit)
 
where D 1  is the delay time through cable  54 , D 2  is the delay time through cable  56 , D 3  is the delay time through cable  57 , D 4  is the delay time through cable  55 , V Transit  is the transit time from the voltage sensing probe  31  to terminal  38  of DUT  40 , and I Transit  is the transit time from the current sensing probe  32  to terminal  38  of DUT  40 , as shown in  FIG. 1 . The signal delays should agree with the above formulas within an error of one-half the sampling interval of the digital scope or other recorder  70 , to produce a valid replica of the DUT waveform for VF-TLP measurement. The specified delays provided by cables  54 - 57  need to be adjustable to the tens of picoseconds. This can be done using constant impedance delay lines that are adjustable in length. The present invention includes providing for the tuning of delay cable lengths to provide more exact timing delays.
 
     FIG. 2  diagrams the output signal from adder  58  (or adder  59 ) that will be recorded by recorder  70  (the bottom trace) relative to the signals that were present at the point where path  20  connects to terminal  38  of DUT  40  (the top trace), and the signals generated by sensing probe  30  (the center trace). 
   The preferred embodiment of the present invention is shown in the detailed schematic of  FIG. 3 . A TLP pulser  100  is shown with two main components: a high voltage supply to charge a transmission line  102  (shown as a 5 nanosecond charge line) and a normally open 50 ohm switch  104  that, when closed, will discharge the energy stored in the transmission line  102  into an output cable  110  as a TLP pulse. For example, if the high voltage supply provides 1000 volts to charge transmission line  103 , when switch  104  closes, a 500 volt pulse having a pulse width of 10 nanoseconds will be coupled to transmission line  110 . There are several variations for generating such pulses known to those skilled in the TLP art, and this is a simple example of one such pulser. 
   To reduce reflections that may cause undesirable multiple pulses, a resistive attenuator  200  is often inserted at some point into the pulse transmission path. The construction of components  110 ,  200 ,  210 ,  300  and  330  of resistive attenuator  200  are designed to maintain a constant impedance. The common impedance is 50 ohms for most high frequency circuits, but other impedances can be used. It is also possible to transform impedances along the pulse path with the use of impedance matching transformers. In a preferred embodiment, all of the components in the pulse measurement circuit should maintain some constant impedance. As is well known in the art, if a component can not be made a constant impedance, it needs to be physically small compared to the highest wavelength of interest. 
   The resistive attenuator  200  reduces reflections, making the source impedance driving cable  210  approximate the cable characteristic impedance, Zo, which is 50 ohms in the preferred embodiment. An exemplary fixed resistive attenuator  200  for reducing the power in pulse reflections, thereby lessening unwanted repetitive pulse stresses, is described in U.S. Pat. No. 6,429,674. Sensing probe  300  is a voltage pickoff probe in this example. Pulse detection with inductive current probes can also be used. Both voltage pickoff probes and inductive current probes may be used as in  FIG. 1 . For clarity, only the voltage pickoff probe has been shown in  FIG. 3 . 
   As seen in  FIG. 3 , voltage sensing probe  31  preferrably comprises a voltage pickoff circuit  300 . Circuit  300  includes a main line input terminal connected to cable  210 , a main line output terminal connected to cable  330  and a signal output terminal connected to cable  410  and thereby to a pulse adder circuit  500 . In the preferred embodiment, voltage pickoff circuit  300  is composed of two resistors. Resistor  320 , with resistance R 320 , forms a resistor divider with the cable  410  impedance. The signal output voltage attenuation of this embodiment equals the main line output voltage times 
               Zo     Zo   +     R   320         =       50     50   +   1450       =     1   30         ,         
where Zo is the cable impedance of  410 . Cables  410 ,  510 ,  550  and  560  are chosen in the preferred embodiment to equal the input impedance of the oscilloscope  600 . The voltage signal output from circuit  300  that is coupled to adder circuit  500  is a replica of the main line voltage reduced by a factor of 30. The voltage pickoff circuit  300  also produces a replica of the reflected pulse. Resistor  310 , with resistance R 310 , is used to match the main line output impedance to cable  330 , which is 50 ohms in this embodiment. If the mainline output impedance is not matched, the reflected pulse returning to the voltage pickoff circuit  300  will be reduced by a re-reflection. Resistance R 310  in this embodiment is selected by calculating
 
             R   310     =         Zo   2       R   320       =       2500   1450     =     1.724   ⁢           ⁢     ohms   .                 
Those familiar with the art of voltage pickoff circuit designs will understand that two matching resistors are commonly used so as to create a symmetrical voltage pickoff. However, this common pickoff design will produce different voltage attenuations for the incident and reflected pulses. In a preferred embodiment of the present invention, by using resistance values for resistors  310  and  320  as described above, the voltage sensing probe is able to measure the incident and reflected pulse voltages with the same attenuation factor.
 
   The signal entering adder circuit  500 , as seen in  FIG. 3 , is split into two identical signals by a splitter  520 , to produce two output pulses that are one-half the voltage of the original signal. This splitter  520  could also be a power divider composed of two 50-ohm resistors. 
   In the preferred embodiment, the splitter  520  and combiner  530  are of similar designs. Between splitter  520  and combiner  530  are pulse delay cables  510  and  550  of different lengths, to thereby create different predetermined pulse delays. Cable  510  is a cable having a delay time that is longer than the pulse width of the TLP pulses being used to test the DUT. In the preferred embodiment, the delay of cable  510  is twice the pulse width of the TLP pulse, which creates a space between the replicated incident pulse signal and the replicated DUT signal. Cable  550  provides a longer pulse delay that is equal to the length of the delay of cable  510  plus twice the pulse delay that exists between voltage pickoff circuit  300  and terminal  380  of DUT  400 , the DUT under test (i.e., the delay provided by cable  550  needs to match the voltage probe to DUT pulse transit time plus the transit time of the reflected pulse back to the probe). As noted above, the time delays of cables  510  and  550  must be precisely set. In a preferred embodiment, cable  550  comprises the series combination of a fixed cable and a constant impedance variable delay line, where the variable delay line preferably has a resolution of 25 picoseconds or less. 
   The two different time-delayed TDR-S signals are added together at the opposite ends of cable  510  and  550  from splitter  520  by combiner  530 . The preferred embodiment uses a matched impedance combiner  530  to add the signals. The output of the combiner  530  equals the sum of the signals from cables  510  and  550 , reduced by a factor of one-half. The result is the signal shown in the bottom waveform of  FIG. 2 . The total attenuation of the signal through the adder circuit  500  is therefore preferably a factor of four. 
   Referring again to  FIG. 2 , the top trace in  FIG. 2  shows pulse # 1 , the waveform at a terminal  380  of DUT  400 . The center trace of  FIG. 2  shows the signal from the voltage pickoff probe  300 . This signal includes the incident pulse # 2  from the pulser and the reflected pulse # 3 . Pulse # 1  at terminal  380  of DUT  400  is the algebraic sum of pulses # 2  and # 3 . The adder circuit  500  of  FIG. 3  processes the measured waveform of the center trace, thereby producing the waveforms shown in the lower trace of  FIG. 2 . As seen in the lower trace of  FIG. 2 , the output of adder circuit  500  includes a pulse # 4 , which is replica of the incident pulse, followed by a pulse # 5 , which is a replica of the pulse at terminal  380  of DUT  400 , further followed by a pulse # 6 , which is a replica of the reflected waveform from terminal  380  of DUT  400 . After the three replica signals, there is a sequence of reflections generated in the adder circuit  500  that are of no consequence. 
   The sequence of pulses including the added signal, or pulse # 5  of  FIG. 2 , is delivered to the digital oscilloscope  600  by cable  560 . This second pulse in this waveform is of special interest as it is a replica of the voltage at the pulsed DUT terminal  380 . In the preferred embodiment, the vertical gain of scope  600  is set so that pulse # 5  is digitized by scope  600  at half of the total voltage range of the scope  600 . This allows the vertical gain and offset to be set to use at least one-half of the scope&#39;s dynamic range in the recording of the replica of the DUT voltage, to thereby provide a low noise measurement. A computer algorithm may be used to set the gain and offset of the scope based on previous analogous measurements. 
   The foregoing descriptions of specific embodiments of the present invention have been presented for purposes of illustration and description. They are not intended to be exhaustive or to limit the invention to the precise forms disclosed, and obviously many modifications and variations are possible in light of the above teaching. The embodiments were chosen and described in order to best explain the principles of the invention and its practical application, to thereby enable others skilled in the art to best utilize the invention and various embodiments with various modifications as are suited to the particular use contemplated. It is intended that the scope of the invention be defined by the claims appended hereto and their equivalents.