Abstract:
A switching circuit has a first Field Effect Transistor (FET) having a first source, a first gate and a first drain, a second FET having a second source coupled to the first source and a second gate coupled to the first gate, a first diode having a first anode coupled to the first source and a first cathode coupled to the first drain, and a second diode having a second anode coupled to the second source and a second cathode coupled to the second drain. In addition, a load is coupled to the switching circuit and a control circuit is coupled to the switching circuit.

Description:
CROSS-REFERENCES TO RELATED APPLICATIONS  
       [0001]     This application is a continuation-in-part of, and claims priority to, co-pending application having Ser. No. 10/763,664 (attorney&#39;s docket number 200300840-1, entitled “Alternating Current Switching Circuit”) which was filed on Jan. 23, 2004. This application is a continuation-in-part of, and claims priority to, co-pending application having Ser. No. 10/764,409 (attorney&#39;s docket number 200311455-1, entitled “Power Converter”) which was filed on Jan. 23, 2004 and which is hereby incorporated by reference herein. 
     
    
     BACKGROUND  
       [0002]     Alternating Current (AC) power control provides a unique set of challenges to those working in the field. There are few solid state electrical devices, such as thyristors and triacs, that will allow AC power to be controlled directly. For both thyristor and triacs the switching times are comparatively long. These long switching times typically limit these devices to low frequency applications, typically AC frequencies of 50-60 Hz. Additionally, full wave rectification to convert AC to direct current (DC), to facilitate work with DC, can result in, among other things, undesirable current harmonics, high frequency conducted emissions that, if not filtered, result in unacceptable noise going back to the power company on the AC power supply lines, and power losses associated with the hardware for performing the full wave rectification.  
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0003]     Embodiments of the present invention will be described by way of exemplary embodiments, but not limitations, illustrated in the accompanying drawings in which like references denote similar elements, and in which:  
         [0004]      FIG. 1  illustrates an AC MOSFET switch, including anti-parallel diodes, in accordance with one embodiment.  
         [0005]      FIG. 2  illustrates a more detailed look at an AC MOSFET switch, including intrinsic parasitic diodes of the MOSFETs, in accordance with one embodiment.  
         [0006]      FIG. 3  illustrates current that is delivered to a load when one embodiment of the AC MOSFET switch is utilized to control current.  
         [0007]      FIGS. 4A-4C  illustrate a power filter and its effects on the current drawn by a load driven by an AC MOSFET switch, in accordance with one embodiment.  
         [0008]      FIG. 5  illustrates an AC MOSFET switch design including a snubbing device, in accordance with one embodiment.  
         [0009]      FIG. 6  illustrates a single IC device containing two NMOS type MOSFET devices of an AC MOSFET switch, in accordance with one embodiment.  
         [0010]      FIG. 7A  illustrates a portion of an embodiment of an inductive heating system utilizing an embodiment of an AC MOSFET switch, in accordance with one embodiment.  
         [0011]      FIG. 7B  illustrates a model for an inductive heating element, in accordance with one embodiment.  
         [0012]      FIG. 8A  illustrates an embodiment of a totem pole configuration of two AC MOSFET switches driving a series resonant circuit, in accordance with one embodiment.  
         [0013]      FIG. 8B  illustrates the timing, for one embodiment, of the two gate drive signals with respect to their reference points as supplied to AC MOSFET switches.  
         [0014]      FIG. 9  illustrates an embodiment of an inductive heating system utilizing an embodiment of an AC MOSFET switch, in accordance with another embodiment.  
         [0015]      FIG. 10  illustrates an embodiment of a subsystem utilizing an embodiment of an AC MOSFET switch to provide power control to a printer fusing system using a resistive type heating element, in accordance with one embodiment.  
         [0016]      FIG. 11  illustrates an embodiment of a subsystem utilizing an embodiment of an AC MOSFET switch to provide power to a single phase, alternating current inductive motor, in accordance with one embodiment.  
         [0017]      FIG. 12A  illustrates a duty ratio of current delivery during startup of an exemplary induction motor, in accordance with one embodiment.  
         [0018]      FIG. 12B  illustrates the current delivered to an embodiment of an AC load over a two second ramp up period corresponding to  FIG. 12A , in accordance with one embodiment.  
         [0019]      FIG. 13  illustrates a duty ratio for operating an embodiment of an AC MOSFET switch during startup, in accordance with another embodiment.  
         [0020]      FIG. 14  illustrates an embodiment of an imaging device, suitable for housing an apparatus utilizing an embodiment of an AC MOSFET switch, in accordance with one embodiment.  
     
    
     DETAILED DESCRIPTION OF THE EMBODIMENTS  
       [0021]     Although specific embodiments will be illustrated and described herein, it will be appreciated by those of ordinary skill in the art that a wide variety of alternate and/or equivalent implementations may be substituted for the specific embodiments shown and described without departing from the scope of the present invention. This application is intended to cover any adaptations or variations of the embodiments discussed herein. Therefore, it is manifestly intended that this invention be limited only by the claims.  
         [0022]     The following discussion is presented in the context of MOSFET devices. It is understood that the principles described herein may apply to other transistor devices.  
         [0023]     Refer now to  FIG. 1  wherein an AC MOSFET switch  110 , including anti-parallel diodes  112   114 , is illustrated, in accordance with one embodiment. For the MOSFETs  142   144  illustrated, the sources of the MOSFET devices are coupled at junction  102 . In one embodiment, MOSFETs  142   144  are power MOSFETs. In addition, the gates are electrically coupled at junction  104 . These couplings are to facilitate the operation of the two MOSFETs  142   144  as a single AC MOSFET switch. Thus, by applying a gate to source voltage, V GS , greater than the threshold voltage, V TH , to the two MOSFETs  142   144 , both MOSFETs conduct current  120 .  
         [0024]     Also illustrated in  FIG. 1  are two diodes  112   114 . These diodes  112   114 , which may be parasitic or explicit, are anti-parallel to their respective MOSFETs. As described in further detail below, these diodes  112   114  may be utilized to bypass the intrinsic anti-parallel diodes of the MOSFETs. Thus, as illustrated, the anodes of the diodes  112   114  are coupled to the sources of the diodes&#39; respective MOSFET and the cathodes are coupled to the respective drains.  
         [0025]      FIG. 1  also illustrates the AC MOSFET switch in use in controlling power to a load. As previously mentioned, AC MOSFET switch  110  comprises two MOSFETs  142   144 . AC MOSFET switch  110  controls current  120  through load  130 . This may be accomplished by switch control circuit  140  which applies the gate-source voltages for the two MOSFETs  142   144  forming the AC MOSFET switch  110 . In the embodiment illustrated, charge pump biasing circuit  150  supplies current to switch control circuit  140  from line (L)  172  and neutral (N)  174  connections of the AC power source.  
         [0026]      FIG. 2  illustrates a more detailed look at an AC MOSFET switch, utilizing P type MOSFETs, including intrinsic parasitic diodes  232   234  of the MOSFETs  242   244 , in accordance with one embodiment. Also illustrated are antiparallel diodes  212   214  which may be utilized to bypass the intrinsic anti-parallel diodes  232   234  of the MOSFETs. Note that the sources of both MOSFETs  242   244  are coupled  204  to each other. In addition, the gates of both MOSFETs  242   244  are coupled  206  to each other. When a voltage, V SG    280  less than a threshold voltage V TH  is applied, the MOSFETs  242   244  will be “turned-off” and the internal reverse biased PN junctions will substantially prevent current from flowing through the MOSFETs.  
         [0027]     When a voltage, V SG    280  greater than a threshold voltage V TH  is applied to the common sources and gates of MOSFETs  242   244  are turned on to facilitate the flow of current through the AC MOSFET switch. Note that current will flow in the reverse direction in MOSFET  242  or  244  depending on the polarity of the AC voltage source. That is, in the reverse direction as is normally used in DC circuits, that is drain to source in an N type MOSFET or source to drain in a P type MOSFET. The reverse current flow causes no problem as the MOSFET transistor is truly a bidirectional device, that is, current may flow from drain to source or source to drain once the proper gate voltage is applied and the conductive channel forms. Normally, during reverse polarity across the source/drain of a MOSFET, an internal PN junction, represented by parasitic diodes  234  and  232  in  FIG. 2 , will eventually turn on allowing current  271  to flow. Note that parasitic diodes  234  and  232  are not separate from the MOSFET  244  and  242 ; e.g. parasitic diode  234  is a PN junction that is part of the structure of transistor  244 . Once the gate voltage is removed the parasitic diode conducts during reverse current flow which makes a single MOSFET unsuitable for the control of alternating current  271   273 . The common source configuration of MOSFET  242  and  244  of  FIG. 2  results in one of the parasitic diodes in a reverse biased state which substantially prevents current flow through the parasitic diodes  232   234  when the MOSFETs are in either the conducting or nonconducting states.  
         [0028]     Referring again to  FIG. 1 , switch control circuit  140  and charge pump circuitry  150  are utilized to provide control for the application of the voltage to the gates of MOSFETs  142   144 . In the embodiment illustrated, switch control circuit  140  may be an externally controlled pulse width modulation circuit. In the embodiment illustrated, charge pump  150  utilizes the AC line to power the pulse width modulation circuitry. In addition, the frequency of the modulated control signal may be fixed, whereas the duty cycle of the modulation, as described below, is utilized to determine the power to be delivered to the load  130 . In an alternative embodiment the gate and source of the AC MOSFET may be driven by a circuit which has a minimum conduction time combined with a variable frequency to determine the power to be delivered to the load  130 .  
         [0029]      FIG. 3  illustrates current that is delivered to a load when one embodiment of the AC MOSFET switch is utilized to control current. For example, as discussed above with respect to  FIG. 1 , the switch control circuit  140  may be a pulse width modulation circuit. In such a case, the power delivered to the load  130  can be controlled by changing the duty cycle of the pulse control signal.  FIG. 3  illustrates an example input voltage  310  from the line and neutral. Illustrated also, in the dark shaded regions  320 , are the periods where the AC MOSFET switch  110  is switched on to allow current to flow through the load  130 . The voltage  310  and current  320  are normalized so that they share a common envelope. Thus, in the illustrated embodiment, a 50% duty cycle signal driving the gate to source voltage will result in an effective power of one half the total power available being delivered the load. By utilizing a pulse width modulation technique, the level of power delivered to the load can be adjusted by controlling the width of the pulses generated by the pulse width modulation of the switch control circuit. The equation governing the power transfer to the load is:  
       Pavg   =         Vrms   2     R     ·     d   .             
 Where V ms  is the Root Mean Square (rms) voltage of the AC power source, R is the resistance of the load and d is the duty ratio of the pulse width modulator driving the AC MOSFET. By inspection of this equation, the power transferred to the load is a linear function of the duty ratio of the pulse width modulator. The load is at zero power when the duty ratio is zero and at maximum power when the duty ratio is 1. 
 
         [0030]     In an alternative embodiment in which the gate and source of the AC MOSFET switch are driven by a circuit which has a minimum conduction time combined with a Variable Frequency Oscillator (VFO) the power delivered to the load  130  is determined by 
 
 P=V   2   ÷R×ƒ×T   min 
 
 Where V is the rms voltage of the AC power source, R is the resistance of the load, f the frequency of the VFO driving the AC MOSFET and T min  the minimum conduction time allowed. By inspection, this equation shows that the power transferred to the load is a linear function of the frequency of the VFO. The load is at zero power when the VFO frequency is 0 and at maximum power when the period of the frequency of the VFO is equal to or less than the minimum allowed conduction time T min . 
 
         [0031]     The above examples operate to facilitate the switching of the alternating current at relatively higher frequencies. There are advantages to switching the current at relatively higher frequencies. Switching frequencies out of the audio range (e.g. greater than 20 KHz) can be utilized to reduce human factor issues associated with audible switching noise. Another advantage of operation at higher frequencies may be a reduction in switching and conduction losses. Implementations operating at significantly lower frequencies spend more time in the linear region of operation. Spending more time in the linear region during switching may dissipate significant amounts of additional energy in the form of heat as relatively slow transitions are made through this linear region. In addition, because of the relatively low voltage drops associated with the disclosed switching of alternating current, less energy is dissipated from the product of the current flowing across the voltage drops of the devices. In addition, the AC MOSFET switching circuit above does not introduce significant harmonics into the alternating current. This can reduce costs associated with filtering these harmonics to meet international regulatory requirements.  
         [0032]      FIG. 4A  illustrates input circuitry for an AC MOSFET switch, in accordance with one embodiment. Illustrated is a filter stage  410  to provides a high frequency short to ground to any transients or conducted emissions that occur across the inputs. Illustrated also is a filtering stage  420  to provide smoothing of the alternating current drawn by the load  430 . The effect of this filter is to smooth the harmonic rich current drawn by the pulse width modulated, or VFO driven load, such that the power source experiences a continuous current flow with virtually no harmonic current content.  
         [0033]     In the embodiment, switch control circuit  450  switches the current  472  delivered to the load as illustrated in  FIG. 4B . During times of switching, assuming a purely resistive load, the current  472  through the load  430  will follow the line voltage provided, that is, it will be in phase. When the switch is turned off, the current delivered to the load will drop to zero  474 . Thus, as can be seen there will be dramatic shifts or steps in the current drawn by the load as the switch turns on and off. These step changes in the current represent unwanted current harmonics placed on the AC power source which may exceed regulatory limits. To solve this problem, filtering stage  420  is added to the circuit.  FIG. 4C  illustrates the current drawn from the AC power source at the line and neutral connections by the switched load as a result of the filtering stage  420 . When the switch is turned off, the filtering stage  420  smoothes current  476  drawn by the load  430 . In the case in which the switch is driven by a pulse width modulator, the total instantaneous current drawn by the circuit may be the sum of the fundamental current and the instantaneous value of the ripple current. This instantaneous current may be expressed as  
           i   L     ⁡     (   t   )       =           V   ·   d     R     ·     sin   ⁡     (     2   ·   π   ·     f   o     ·   t     )         +         π   2     4     ·     (     1   -   d     )     ·       (       f   c       f   s       )     2     ·       V   ·   d     R     ·     sin   ⁡     (     2   ·   π   ·     f   o     ·   t     )       ·       sin   ⁡     (     2   ·   π   ·     f   s     ·   t     )       .             
 
 where f c  is the resonant frequency of filtering stage  420 , f s  is the switch frequency of the pulse width modulator, f o  is the frequency of the AC power source, d is the duty cycle of the pulse width modulator, V is the peak source voltage, and R is the load resistance  430 . Under direct examination of this equation it is noted that, as the switch frequency of the pulse width modulator is increased, the resultant alternating current waveform at the Line and Neutral connections smoothes dramatically. 
 
         [0034]      FIG. 5  illustrates an AC MOSFET switch design including a snubbing device  580 , in accordance with one embodiment. Snubbing device  580  is utilized for dissipating energy stored in the circuit. Stored energy in a circuit exists due to various factors associated with the circuit such as: parasitic inductance associated with the wiring providing the AC current, parasitic inductance in the components leads, and inductance in the load itself. Snubber designs are designed to capture a portion of the stored energy in a circuit, when the circuit is switched off. These snubber designs are to reduce, among other things, the resonance of the circuit. However, these snubber designs are not engineered to dissipate all the energy; they are simply designed to dissipate enough energy to reduce resonance and the resulting resonant “over” voltages that may otherwise occur.  
         [0035]     To dissipate all the energy in the circuit, a significantly larged sized capacitor  573  may be used in snubber  580  design. It is desirable to have the resistance  577  approximately match the resistance in the load  530 . Thus, if the load resistance is approximately 20 ohms, then the resistance of the snubber should be selected to be about 20 ohms. In addition, the stored inductance  575  for a typical circuit driving the AC MOSFET switch has been measured at approximately 100 nanoHenries. In some snubber designs, a capacitor capable of capturing about ⅕ of the energy stored in the inductive parasitics may be utilized. As mentioned, this capacitor size is utilized to simply avoid resonance of the circuit. However, the remaining energy is dissipated via heat in the switching element or as Radio Frequency (RF) emissions. To avoid this heat or RF emissions, a larger snubber circuit may be utilized.  
         [0036]     In order to have the snubber dissipate substantially all the stored energy of the circuit, the energy dissipated by the snubber should equal the energy stored due to the inductance of the circuit. Thus, 
 
½ LI   2 =½  CV   2 , where  I=V/R 
 
½  L ( V/R ) 2 =½  CV   2 
 
 Solving for C we find that: 
 
 C=L/R   2 
 
 Thus, the capacitor used is directly related to the value of the parasitic inductance. 
 
         [0037]     Dissipating heat may be undesirable as it may result in damage to the circuit. A solution to this may be to include a heat sink. However, the addition of the heat sink may add cost to the design. In addition, generation of RF emissions may be undesirable as it may result in poor classification during RF certification proceedings for the device containing the AC MOSFET switch. To protect from RF emissions, a shield for the RF emissions may be provided. Again, however, the addition of a shield may add cost to the design.  
         [0038]     Thus, in one embodiment, the capacitor that is part of the snubber illustrated in  FIG. 5  is designed to capture substantially all of the stored energy in the circuit associated with the AC MOSFET switch. In this manner, the design of RF shield and the design of any heat dissipating devices may be reduced.  
         [0039]      FIG. 6  illustrates a single integrated circuit (IC) device  600  containing two NMOS type MOSFET devices of an AC MOSFET switch, in accordance with one embodiment. In an alternative embodiment, two PMOS type MOSFET devices may be utilized in the construction of an AC MOSFET switch. Recall that the two sources from the two MOSFETs are logically coupled to each other in the AC MOSFET switch. By fabricating the two MOSFETs in a single package on an IC, the two MOSFETs may share a common source region  610  on the IC. In the embodiment illustrated in  FIG. 6 , a common source region  610  is implanted into the die containing the AC MOSFET switch. The sharing of the common source region  610  may allow the use of a single source lead emanating from the package containing the two MOSFETs of AC MOSFET switch. This, in turn, may result in decreased conduction resistance due to the elimination of one source lead and the source lead&#39;s associated wire bonding parasitics, such as ohmic resistance from the die to a package lead. For example, in one embodiment, the elimination of one of the source leads may reduce the impedance by 70 milliohms, corresponding to the impedance associated with one of the leads to the AC MOSFET switch.  
         [0040]     70 milliohms may be a substantial portion of the overall resistance associated with the AC MOSFET switch. For example, assume an R DSON  of 100 milliohms for each MOSFET in the AC MOSFET switch. Thus, with a 70 milliohm resistance for each lead for the source and drain, the overall path impedance across the source and drain is 240 milliohms. Two discrete series devices have an effective resistance through the AC MOSFET switch of 480 milliohms. Recall that the external source lead in the AC MOSFET is used for the application of gate bias and as a conduction path for certain types of snubber applications during switch turn off. By design the external source connection  610  has very low current flow and does not introduce series resistance to the AC MOSFET switch when the switch is conducting. This fact allows the conduction resistance of the AC MOSFET switch to be reduced by 140 milliohms, or a reduction in effective resistance 30% by using a common source region on the die of the AC MOSFET and the elimination of one lead. Since the power dissipated is directly related to the resistance, this results in a 15% reduction in power loss, for the embodiment described. Fabrication of the AC MOSFET switch on a single die also allows one of the gate terminals of the discrete implementation to be eliminated. The result of the common source region and eliminated gate terminal is a four pin device with two high current drain connections and two lower current gate and source connections. One pin of the four pin device is coupled to each of the gates of the two MOSFETs. Another pin is coupled to the common source region , and each of the two remaining pins are coupled to a different one of the drains.  
         [0041]     The AC MOSFET switch may be utilized in various devices and/or systems to control AC loads, in particular, inductive loads. Examples of systems with inductive loads include but are not limited to subsystems of photocopier and laser printing systems. Such subsystems may include fuser power control subsystems and inductive heating subsystems. Other devices, such as home appliances, containing induction motors may also utilize AC MOSFET switches for AC power control.  
         [0042]     In the figures that follow, various aspects of the details of the AC MOSFET switch, such as the antiparallel diodes, are occasionally omitted to simplify the figures in order to not obscure the embodiments being described.  
         [0043]      FIG. 7A  illustrates a portion of an inductive heating system utilizing an embodiment of an AC MOSFET switch  710 , in accordance with one embodiment. Utilizing the AC MOSFET switch  710  to control power to an inductive heating element  720  may reduce a substantial amount of the device losses, and possibly, as much as halves the total switching losses. In one embodiment, assuming a drain to source conduction resistance of 0.07 ohms in each of the two MOSFETs, the power dissipated is approximately: 
 2*0.07*8*8=8.96 Watts 
         [0044]     As illustrated in FIG.,  7 B, inductive heating element  720  may be modeled as a simple N:1 transformer with a single turn on the secondary winding which is then connected to a very low value resistive load capable of handling very high power loads. Temperature sensor  730  may be utilized to provide a measurement of the heating element&#39;s temperature to the control circuit  740 . Temperature sensor represents a typical temperature sensor, such as a thermistor, and will not be described further. The control circuit  740  may be utilized to provide control for AC MOSFET switch  710 . That is, the control circuit may be utilized to determine when to allow alternating current to flow through the inductive heating element  720 , thus controlling the power to the inductive heating element  720 . An example of a control circuit  740  suitable for use with AC MOSFET switch  710  in controlling power in an inductive heating system is the control circuit disclosed in U.S. Pat. No. 5,789,723 titled “Reduced Flicker Fusing System for Use in Electrophotographic Printers and Copiers” (herein incorporated by reference). In alternate embodiments, other equivalent control circuits may be employed instead.  
         [0045]     Bias circuitry  750  may be utilized to bias the control circuitry  740  and provide reference voltage for the gate to source voltage utilized in the biasing of the AC MOSFET switch  710 . An example of a biasing circuit  750  suitable for use with the novel AC MOSFET switch  710  is the biasing circuit disclosed in U.S. Pat. No. 6,396,724 titled “Charge-pumped DC Bias Supply”. In alternate embodiments, other equivalent biasing circuits may be employed instead.  
         [0046]     Recall that the AC MOSFET switch biasing voltages across the gate/source can float with respect to the voltage applied across the AC MOSFET switch  710 . Accordingly, the biasing circuit  750  may be employed to electrically decouple or isolate the control circuit  740  from the AC power circuit. This may be performed using an isolation transformer. Note, however, that while using a transformer to provide isolation provides galvanic isolation, non-galvanic isolation is also possible; as long as the bias circuit can float with respect to the line or neutral.  
         [0047]     R S C S    760  form a turn-off snubber for the AC MOSFET switch. Thus, upon switching the current off at the AC MOSFET switch  710 , the energy stored in the parasitic inductance of the circuit can be dissipated through resistor/capacitor combination, instead of being directed at, and dissipated by, the AC MOSFET switch  710 .  
         [0048]      FIG. 8A  illustrates an embodiment of a totem pole configuration of two AC MOSFET switches driving a series resonant circuit, in accordance with one embodiment. The series resonant circuit comprises an induction coil  826  and capacitor C  827 . The series resonant circuit is used to heat a fusing unit in a laser printer and only the primary resonant circuit is shown. This totem pole configuration provides an ability to handle high resonant currents resulting from the switching off of an inductive load such as that of the induction coil  826 . In this embodiment the power transfer from the line L and neutral N terminals of the AC power source to the item undergoing induction heating is a linear function of the drive frequency applied to the AC MOSFET switches as given in the following equation:  
       P   ∝       1   2     ⁢     f   ·   C   ·     V   2               
 where f is the switch drive frequency, C the value of the series capacitance  827  and V the rms voltage of the AC power source. The totem pole configuration comprises two back-to-back AC MOSFET switches  822   824 . In the embodiment illustrated, each of the two AC MOSFET switches  822   824  are controlled by control circuit  810 . By utilizing two AC MOSFET switches  822   824  higher resonant currents may be tolerated. 
 
         [0049]      FIG. 8B  illustrates the timing, for one embodiment, of the two gate drive signals  801   803  with respect to their reference points  802   804  as supplied to AC MOSFET switches  822   824 . Reference point  802  is attached to the common source of AC MOSFET  822  and reference point  804  is attached to the common source of AC MOSFET  824 .  FIG. 8B  also illustrates the resulting current waveform  828  through the induction coil  826  with respect to the AC MOSFET drive signals. Assuming that AC MOSFET  822  is conducting and the voltage at the L (line) terminal is positive and the voltage at the N (neutral) terminal is negative, current waveform  828  will flow into coil  826  charging capacitor  827  and continue to the N terminal completing the circuit. This current flow starts out at zero amperes and will climb to a maximum and then attempt to resonate. The effective resistance across the secondary winding of the induction coil results in a highly damped oscillatory circuit and the resonant oscillations of current waveform  828  quickly die away to zero amperes. During this resonant current flow any metallic device placed near induction coil  826  may experience induced currents which result in energy transfer and the desired resulting heating of the metallic device placed near the induction coil  826 . Next AC MOSFET  822  is turned off and, after a small time delay  867 , AC MOSFET  824  will start to conduct. The time delay  867 , in which both AC MOSFETs are off, is many times referred to as “dead time” or as a “blanking interval”. This time delay  867  may be utilized such that neither AC MOSFET will conduct at the same time. Such concurrent conducting may result in an effective ‘short circuit’ across the AC power source which may result in the destruction of the two AC MOSFETs  822   824 . Next AC MOSFET  824  starts to conduct and a current will start a reverse flow out of capacitor  827  through induction coil  826  proceeding through AC MOSFET  824  and completing the circuit at capacitor  827 . Energy is again transferred via the induction coil to the metallic device to be heated. The resonant current quickly dies out, and then AC MOSFET  824  is turned off and another period of dead time is applied. The process then repeats. Switching the AC MOSFETs  822   824  on and off at zero current substantially reduces the typical losses experienced while switching an inductive circuit and result in a significant increase in the efficiency of the converter.  
         [0050]      FIG. 9  illustrates an embodiment of an inductive heating system utilizing an embodiment of an AC MOSFET switch  910 , in accordance with another embodiment. In this embodiment, to increase converter efficiency by reducing converter losses, a Cuk topology may be applied to an AC MOSFET switch design for an inductive heating element. When the AC MOSFET switch  910  is on, current builds in L 1    920 . In addition, charge from C 1    930  is transferred through inductive coil L C    940  to the load. This transfer of energy via an inductive heating element can be simply modeled as an N:1 transformer with a resistive load on the secondary winding. When the AC MOSFET switch  910  turns off, the energy in L 1    920  is forced into the series resonant C 1 L C  load. This approach reduces turn off switching loss in the AC MOSFET switch  910 .  
         [0051]      FIG. 10  illustrates an embodiment of a subsystem utilizing an AC MOSFET switch  1005  to provide power control to a printer fusing system using a resistive type heating element  1040 , in accordance with one embodiment. L 1 C 1    1010   1012  act as a low-pass filter such that the AC source at L  1015  and N  1017  experiences a near pure resistive load for practical power levels. For example, a load ranging between 10 Watts to 1000 Watts enjoys a near unity power factor over the entire load range. R 1    1013  is a safety precaution to discharge C 1    1012  when the power controller of  FIG. 10  is disconnected from the AC power source such that the voltage appearing at the L  1015  and N  1017  terminals is quickly reduced to zero volts. Capacitor C 0    1011  may be placed across the L  1015  and N  1017  terminals to filter out conducted emissions that may otherwise be injected into the AC power source. The value specified for C 0    1011  may be determined empirically. For example the value may be such that the power converter meets regulatory standards. However, the value typically falls into a range of approximately 1 uF per kilowatt that the converter controls. Thus, for a 1.2 kW maximum power load a standard value capacitance of 1.47 uF would be chosen for C 0    1011 .  
         [0052]     R 2 C 2    1020   1022  and R 3 C 3    1030   1033  may act as turn off snubbers for the AC MOSFET switch  1005  and the fuser heating element  1040 , respectively, to reduce radiated and conducted emissions. Temperature sensor  1045  may be utilized to monitor the fuser temperature and provide the sensed temperature as feedback to the control circuit  1050 . The control circuit  1050  may be used to control the AC MOSFET switch  1005  and thus to control the current to the fuser&#39;s resistive heating element  1040 . In this instance, resistive fuser heating element, is understood to include, various types of resistive elements such as screen printed film resistors, resistive element heating lamps, open air metallic resistance coils, etc. In one embodiment, the control circuit  1050  comprises a pulse width modulated (PWM) control circuit. In another embodiment, the control circuit  1050  comprises a variable frequency drive that yields a power transfer characteristic that varies with drive frequency. An example of a control circuit  1050  which may be utilized in conjunction with the novel AC MOSFET switch  1005  is the linear control circuit disclosed in U.S. Pat. No. 5,811,764 titled “Method for Reducing Flicker in Electrophotographics Printers and Copiers” (herein incorporated by reference). In alternate embodiments, other equivalent linear control circuits may be employed.  
         [0053]     Bias circuit  1060  may be employed to provide DC voltages and currents for control circuit  1050  as previous discussed. An example bias circuit  1060  is disclosed in U.S. Pat. No. 6,396,724 titled “Charge-pumped DC bias supply” (herein incorporated by reference). Additional examples may be found in U.S. Pat. No. 6,563,726 titled “Synchronous bridge rectifier” (herein incorporated by reference). Finally, a regenerative snubber to provide bias to control circuit utilizing recaptured energy is disclosed in co-pending application Ser. No. 10/780,927 (attorney&#39;s docket number 200309715-1, entitled “SNUBBER CIRCUIT”) filed on Feb. 17, 2004.  
         [0054]      FIG. 11  illustrates an embodiment of a subsystem utilizing an AC MOSFET switch  1110  to provide power to a single phase, alternating current induction motor  1120 , in accordance with one embodiment. L 1    1130  and C 1    1135  form a series resonant low pass filter that filters the current pulses supplied to the motor so that the AC source experiences a continuous load with very low levels of harmonics. R 1    1137  may be employed as a safety feature to discharge C 1    1135  in the event that service to the circuit may be performed. Free wheeling capacitor C 3    1150  provides a continuous path for the current flowing through the motor when the AC MOSFET switch  1110  is turned off while the pulse width modulated control circuit  1160  is active. L 2    1170  acts to limit current through C 3    1150  when the AC MOSFET turns on. Resistor R 2    1142  and capacitor C 2    1140  form a turn off snubber to protect the AC MOSFET switch  1110  from energy stored in the inductance in the system which may cause over voltages in the AC MOSFET switch  1110  when the AC MOSFET switch  1110  is turned off. R 2    1142  and C 2    1140  may also act to reduce the power losses in the MOSFET devices of the AC MOSFET switch  1110 . Additionally, R 2    1142  and capacitor C 2    1140  may filter high frequency electromagnetic noise that may otherwise appear as radiated electromagnetic radiation or conducted radiation which may be injected into the AC power source.  
         [0055]     Another advantage of the utilization of an AC MOSFET switch design with the inductive loads, such as motors, disclosed herein, may be the ability to provide soft start functionality. As a motor begins to spin up after being turned on, the motor can produce a current surge that is approximately five times larger than the maximum rated current for a device. (And, please do not include  FIG. 13  in the filed application.) This large start-up transient may react with the impedance in the wiring from the AC power source and cause voltage sags and surges. These voltage fluctuations may then cause the light output from light sources connected to the AC power source to flicker. Flicker is a very undesirable artifact and much attention is devoted to reducing it in AC power systems. An AC MOSFET switch circuit controlling such an inductive motor may be operated in a manner to facilitate control of this start up current.  
         [0056]      FIG. 12A  illustrates a duty ratio of current delivery during startup of an exemplary induction motor, in accordance with one embodiment. The duty ratio is the ratio of time where switch is turned on versus the total period that it could be on.  FIG. 12A  illustrates a linear duty ratio operation of the AC MOSFET switch by a control circuit to supply current to the inductive load from time  0  to the end of a two second startup period  1220 . Note that the amount of time for a start up period may vary. For a circuit, the start up period may depend upon the characteristics of the AC inductive load in that circuit. For example for some inductive motors, five seconds may be desirable for the startup period. Such a linear duty ration operation of the AC MOSFET switch may be accomplished by utilizing a pulse wave modulation scheme in the control circuit.  FIG. 12B  illustrates the current delivered to an embodiment of an AC load over a two second duty ratio ramp up period corresponding to  FIG. 12A .  
         [0057]      FIG. 13  illustrates a duty ratio for operating an embodiment of an AC MOSFET switch during startup, in accordance with another embodiment. As illustrated, the duty ratio may begin at a 0.2 duty ratio  1310  instead of 0. This initial non-zero duty ratio start is to provide better startup characteristics for an AC load. For example, in the case of an AC induction motor, a quicker spin up of the motor may be obtained by using an initial 0.2 duty ratio. The remainder of the duty ratio illustrates a linear increase until the duty ratio approaches 1 at the end of a five second ramp up period. The time period of the ramp may be chosen to provide the desired motor starting characteristics along with the desired motor current profile, in this embodiment five seconds is the ramp period. In other embodiments different time periods for the ramp may be utilized. While there may still be an initial transient as a result of this initial step, the transient will be considerable smaller then the 5-times rated current surge that occurs with no soft start functionality.  
         [0058]     While the soft start methods are discussed with respect to an induction motor, the techniques apply to any load driven by the AC MOSFET switch, such as the resistive heating element in a printer fusing system or induction heating elements previously described. In addition, while a linear ramp of the current is utilized, one skilled in the art will recognize that other, non-linear ramping of the current to the load may be obtained using the AC MOSFET switch. Further, a similar, but inverse, current ramping can be utilized during the load turn-off to provide further advantages during the power down of the AC load. For example, during the turn-off of certain AC loads, flickering can occur on adjacent incandescent and fluorescent lighting systems. A ramped turn-off of the load driven by the AC MOSFET alleviates this problem.  
         [0059]      FIG. 14  illustrates an embodiment of an imaging system  1400 , suitable for housing an apparatus utilizing an embodiment of an AC MOSFET switch driving an AC load, in accordance with one embodiment. As illustrated, for the embodiment, imaging system  1400  includes processor/controller  1402 , memory  1404 , imaging engine  1406  and communication interface  1408  coupled to each other via bus  1410 . Imaging engine  1406  comprises a fusing subsystem  1420  for fusing toner to paper. In addition to fusing subsystem, imaging system may comprise other inductive heating elements or induction motors. Imaging engine  1406  is similar to those found in many imaging systems, such as those available from Hewlett Packard Corp. of Palo Alto, Calif. Fusing subsystem  1420  is connected to an alternating current power source through interface  1430 .  
         [0060]     Processor  1402 , in combination with other portions of the imaging system  1400 , can perform various control functions of the fusing subsystem  1420 . For example, in one embodiment, processor  1402  controls power management of the fusing subsystem  1420  to intelligently power down the fusing subsystem when the fuser is not in use. Otherwise, processor  1402 , memory  1404 , imaging engine  1406 , comm. interfaces  1408 , and bus  1410  represent a broad range of such elements.  
         [0061]     In various embodiments, imaging device  1400  may be an inkjet printer or an electrophotographic printer.  
         [0062]     Thus, various embodiments are illustrated utilizing an AC MOSFET switch in a circuit delivering current to a load, including an inductive load. Although specific embodiments have been illustrated and described herein, it will be appreciated by those of ordinary skill in the art that a variety of alternative and/or equivalent embodiments may be substituted for those disclosed herein without departing from the spirit and scope of this disclosure. This application is intended to cover any adaptations or variations of the embodiments discussed herein. Therefore it is intended that the present invention be limited only by the claims and the equivalents thereof.