Abstract:
A reception device for detecting an electromagnetic radio-frequency wave emitted by the examination subject in a magnetic resonance tomography installation has two reception coils and amplifiers arranged following these coils. The mutually phase-delayed reception signals of the reception coils are tapped by a common signal line. For compensating the phase shift one of the amplifiers is or both amplifiers are directed between the two reception signals. A separate combiner can thereby be advantageously eliminated.

Description:
BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The present invention is directed to a reception device for detecting an electromagnetic radio-frequency wave emitted by the examination subject of a nuclear magnetic resonance tomography installation, of the type having a first reception coil, a first amplifier allocated to the first reception coil, a second reception coil whose reception signal is phase-shifted with respect to the reception signal of the first reception coil, particularly by a phase angle of 90°, comprising a second amplifier allocated to the second reception coil, and a common signal line via which the output signals of the two amplifiers are carried off. 
   2. Description of the Prior Art 
   For receiving the circularly polarized radio-frequency signal, two orthogonal polarization components are separately received in a magnetic resonance tomography installation. In, for example, a vertical field apparatus, a first polarization component is received parallel to the patient axis and a second polarization component is received perpendicular to the patient axis. For receiving the first polarization component parallel to the patient axis, a loop coil can be used that surrounds the entire patient body or an extremity to be examined. For the reception of the second polarization component, for example, a butterfly coil or saddle coil is used. Both field components of the circularly polarized magnetic field part of the electromagnetic radio-frequency signal can be detected with the separately existing reception coils or reception antennas. 
   The reception signals of the separately existing reception antennas are amplified in separate pre-amplifiers allocated to the reception antennas and are supplied via a 90° coupler (90° combiner) that takes the phase shift between the two polarization components into consideration to a common signal line. Via the common signal line, the radio-frequency signals combined in this way are supplied to the image evaluation, image reconstruction and image display. The phase shift between the two polarization components typically amounts to 90°. However, dependent on the structure of the magnetic resonance tomography installation, other values below this and above this are possible. 
   The combiner is composed of a capacitive impedance and of an inductive impedance. The output impedance of the combiner must be matched to the impedance of the common signal line. The lines leading to the combiner as well as the combiner itself must be shielded in an involved way. The known combiner also has the disadvantage that it requires a certain spatial volume for the circuitry, leading to unergonomically bulky arrangements, particularly in the manufacture of non-stationary, i.e. portable coil arrangements, for example in the manufacture of surface coils, because the combiner must be accommodated on the coil arrangement. 
   SUMMARY OF THE INVENTION 
   An object of the present invention is to provide a reception device for a magnetic resonance tomography installation wherein the aforementioned disadvantages are avoided. 
   This object is achieved in a reception device of the type initially described, wherein at least one of the amplifiers compensates the phase shift between the two reception signals. 
   The invention is based on the perception that the function of the previously employed combiner and be at least partially displaced into the amplifiers or pre-amplifiers, and the combiner can thus be partially or entirely eliminated. 
   The invention achieves the advantage that the reception device can be constructed more simply, smaller, more economically, less susceptible to malfunction and in a more space-saving way. Moreover, fewer adjustment tasks are needed for the impedance matching, so that the manufacturer is also simplified and the testing costs are reduced. The amplifiers and pre-amplifiers, for example, can have a separate or a common shielding housing wherein the function of the previously employed combiner is relocated. 
   The output impedances of the two amplifiers are preferably established between the two reception signals for compensating the phase shift. 
   In particular, a coil at the output side or a capacitor at the output side is used for this purpose. 
   For processing the reception signals that may be of different strength under certain circumstances, it can be necessary to weight these differently. To this end, for example, the gains of the allocated amplifiers can be set differently by connecting active components. Alternatively or additionally, a weighting is undertaken by a suitable setting of the output impedances, particularly of the coil present at the output side and/or of the capacitor present at the output side. It is not only a phase angle but also an amount, namely that can be set via the output impedance. The case of identical weighting by the output impedances, i.e. the case of equal amounts, is referred to below as the “symmetrical case”. 
   In a preferred embodiment, the output impedances of the two amplifiers are fashioned for generating mutually opposite phase angles, particularly for generating phase angles of −45° or +45°. In this embodiment, the amplifiers contribute in a way that is the same in magnitude but oppositely directed to generating the desired phase shift. This is preferably true in the “symmetrical case”. Otherwise, the phase angles can have the same operational sign under certain circumstances, but the difference is adapted to the phase shift between the reception signals that is to be balanced. 
   It is expedient for avoiding reflections at the output of the two amplifiers for the output impedances of the two amplifiers to be additionally directed to the impedance of the common signal line for adaptation. In complex notation, this means that the output impedances of the two amplifiers comprise complex values that accomplish the generation of a phase shift as well as matching to the signal line. 
   Preferably, the output impedances of the two amplifiers—in complex notation—are complex conjugates (of each other), particularly in the symmetrical case. 
   For example, the output impedance of one of the two amplifiers—in complex notation—exhibits the value X+i·X, and the output impedance of the other amplifier exhibits the value X−i·X, whereby X indicates the desired impedance of the common signal line and preferably exhibits a value of 50 Ω. This is preferably true in the symmetrical case. 
   As already mentioned, it is advantageous in the reception device of the invention that a separate combiner as well as an element for impedance matching of the pre-amplifiers to the common signal line can be eliminated. Preferably, the output lines of the two amplifiers are therefore connected to the common signal line either directly, or without interposition of phase-delaying elements, particularly without longer signal paths or lines that produce noticeable phase shifts. 
   Due to the susceptibility to malfunction and shielding, it is expedient for the two amplifiers to be fashioned as an assembly unit and, in particular, to be arranged in a common housing or shielded housing. 

   
     DESCRIPTION OF THE DRAWINGS 
       FIG. 1  illustrates a reception device of the prior art. 
       FIG. 2  is an equivalent circuit diagram for the output of an amplifier. 
       FIG. 3  illustrates a reception device of the invention. 
       FIG. 4  is an equivalent circuit diagram for the output of an amplifier. 
   

   DESCRIPTION OF THE PREFERRED EMBODIMENTS 
   For easier comprehension, a phase shift of exactly 90° between the received polarization components forms the basis of the discussion below. The considerations can be transferred to cases having a different phase shift. 
     FIG. 1  shows a reception device  1  of a magnetic resonance tomography installation whose other component parts are not explicitly shown. The reception device  1  has a first reception coil  3  that is fashioned as a butterfly element. A second reception coil  5  fashioned as a loop element. Matching networks  7  and  9  are respectively integrated into the coil elements  3 ,  5 , these serving the purpose of noise matching. The matching network  7  of the first reception coil  3  is connected via a line  11  to a first pre-amplifier or to a first amplifier  15 . In the same way, the matching network  9  of the second reception coil  5  is in communication via a line  13  with a second pre-amplifier or a second amplifier  17 . 
   The output signals of the two amplifiers  15 ,  17 —respectively shielded with separate housings—are supplied to a 90° combiner  25  via shielded coaxial lines  21  and  23  in the known reception device  1  of FIG.  1 . The 90° combiner  25  has to be separately shielded. At its output, the 90° combiner  25  is in communication with a common signal line  19  implemented as a shielded coaxial cable and via which the signals are supplied in common to the image evaluation, circuitry and other components. 
   The circuit structure of the two amplifiers  15 ,  17  is only schematically shown, without detail. The circuit includes a field effect transistor and a transistor at the output side. For matching the output impedance Z 2,0  of the first amplifier  15  to an ohmic line impedance of R 0 =50 Ω of the line  21 , the amplifier  15  contains a capacitor C 2,0  connected in parallel, and an inductance L 2,0  connected in series, at its output. For matching the output impedance Z 1,0  of the second amplifier  17  to an ohmic line impedance R 0 =50 Ω of the line  23 , the amplifier  17  contains a capacitor C 1,0  connected in parallel, and an inductance L 1,0  connected in series, at its output. 
   Conventionally, the output impedances Z 2,0  and Z 1,0  of the amplifiers  15 ,  17  are respectively matched to a value of R 0 =50 Ω with the capacitors C 1,0  C 2,0  and coils L 1,0 , L 2,0 . The output resistance R of the transistor must be taken into consideration. The following matching condition derives: 
                 Z     2   ,   0       ≡       1       1   R     +     ⅈω   ·     C     2   ,   0             +     ⅈω   ·     L     2   ,   0             =     R   0             Equation   ⁢           ⁢   1             
 
   The above matching condition is reproduced for the first amplifier  15  and analogously applies to the second amplifier  17  with the values C 1,0 , L 1,0 . the radian frequency ω=2·π·f of the magnetic tomograph (for example, 63.6 MHz for a 1.5 Tesla system). The creation of the matching condition is illustrated in detail in the equivalent circuit diagram of FIG.  2 . In  FIG. 2 , the output of the transistor is symbolized by an output resistance R (≈2 kΩ) that is followed by the capacitors C 1,0  or C 2,0  and coils L 1,0  or L 2,0  provided for the matching, and with which a matching to 50 Ω is undertaken. 
   The following dimensioning rules for the capacitor C 2 , 0  derive from the matching condition according to Equation 1: 
               C     2   ,   0       =             R   ·     R   0       -     R   0   2           ω   ·   R   ·     R   0         =     7.8   ⁢           ⁢   pF               Equation   ⁢           ⁢   2             
 
and, for the coil L 2,0 :
 
 L   2,0   =R   0   ·R·C   2,0 =781.4 nH  Equation 3
 
   The combiner  25  shown in  FIG. 1  is reproduced with a schematic circuit diagram. The combiner  25  is composed of a capacitive impedance ZC and an inductive impedance Z L . The output signal V out  is composed—according to the following equation, of the respective input signals V in1  or V in2  of the amplifiers  15 ,  17 : 
                     V   out     =       ⁢             Z   L     +     Z     2   ,   0             Z     1   ,   0       +     Z   C     +     Z   L     +     Z     2   ,   0           ⁢     V   in2       +                     ⁢           Z   L     +     Z     1   ,   0             Z     1   ,   0       +     Z   C     +     Z   L     +     Z     2   ,   0           ⁢     V   in1                     Equation   ⁢           ⁢   4             
 
   In order to realize a 90° combiner, the impedances ZC=−i 50 Ω and ZL=+i 50 Ω must apply given a source impedances Z 1 , 0 =Z 2 , 0 =50 Ω at the input side. The output signal is then established by: 
               V   out     =       1     2       ⁢     (         ⅇ     ⅈ   ⁢     π   4         ·     V   in2       +       e       -   ⅈ     ⁢     π   4         ·     V   in1         )               Equation   ⁢           ⁢   5             
 
   In order to achieve an exact phase difference, the amplifier outputs are usually set to 50 Ω. Trimming capacitors T 1 , T 2  with which the combiner  25  can also be exactly set are provided for the two impedances Z C  and Z L . 
   The reception device  1  of the invention in  FIG. 3 , has a first pre-amplifier or first amplifier  35  and a second pre-amplifier or second amplifier  37  that are dimensioned differently at the output side than are the amplifiers  15 ,  17  of FIG.  1 . 
   Since the two amplifiers  35 ,  37  of  FIG. 3  are tuned relative to one another for achieving the desired phase shift, the combination of the two amplifiers  35 ,  37  can be interpreted as a double-amplifier unit  39  and/or—on the basis of corresponding structural measures, can be constructed as an assembly unit, for example on a shared motherboard. The double-amplifier unit  39  preferably has a common shielding housing, having two inputs and one output in the example. 
   In contrast to the known arrangement shown in  FIG. 1 , the combiner  25  and, in particular, the impedances Z C  and Z L  are superfluous as separate components in the reception device  1  of the invention—which is explained in greater detail in FIG.  3 —because their function is at least partially integrated into the amplifiers  35 ,  37 . This by virtue of the output impedances Z 1  and Z 2  of the two amplifiers  35  and  37 —in contrast to the amplifiers  15 ,  17  of FIG.  1 —being set to a value of 50 Ω−i 50 Ω and 50 Ω+i 50 Ω, respectively. The equivalent circuit diagram of  FIG. 4  be used again for explanation. The matching condition for the symmetrical case accordingly, for example for the first amplifier  35 , is as follows: 
                 Z   2     ≡       1       1   R     +     ⅈω   ·     C   2           +     ⅈω   ·     L   2           =       R   0     +       i   ·   50     ⁢   Ω               Equation   ⁢           ⁢   6             
 
   The following dimensioning rules derive therefrom for the capacitors C 1 , C 2  and coils L 1 , L 2  present at the output side in the amplifiers  15 ,  17  (only shown here for the first amplifier  15 ): 
     C 2 =C 2,0   Equation 7                L   2     =       L     2   ,   0       =         50   ⁢   Ω     ω     =     906.5   ⁢           ⁢   nH                 Equation   ⁢           ⁢   8               
   Thus, due to an increase to 906.5 nH, or a reduction (to 656.3 nH) in the values of the inductances L 2 , L 1  compared to the corresponding values L 2,0 , L 1,0  of the case shown in  FIG. 1 , a phase shift can already realized in the amplifiers  35  or  37 . As a result, it is possible to achieve a desired effect of a combiner by simply connecting the outputs of the amplifiers  35 ,  37  via output lines  41 ,  43  and via a simple T-element or a T-shaped line  45 , namely connection thereof to the common signal line  19 . At the same time, the amplifiers  35 ,  37  are matched to the line impedance Z 0  of the common signal line  19 . The output lines  41 ,  43  can be realized on the circuit board of the amplifiers  35 ,  37  and are expediently only a few mm long (shown longer in  FIG. 3  for purpose of illustration). 
   The amplifiers  35 ,  37  with “quasi-integrated combiner” have the advantage that a separate combiner is not required and, thus, costs for material and testing can be eliminated. A coil module with one or both coils  3 ,  5  and one or both amplifiers  35  and  37  can be constructed smaller as a result. In particular, an array coil can thus be constructed in an especially simple way. 
   Although modifications and changes may be suggested by those skilled in the art, it is the intention of the inventor to embody within the patent warranted hereon all changes and modifications as reasonably and properly come within the scope of the inventor&#39;s contribution to the art.