Abstract:
A modulator apparatus operating at a low supply voltage, configured for receiving an input-voltage signal in base band and supplying an output-voltage signal at a given modulation frequency under control of a signal generated by a local oscillator and comprising a transconductor stage that carries out a voltage-to-current conversion of said input-voltage signal. A voltage-to-current conversion module is coupled to a current-mirror module configured for mirroring a current in a Gilbert-cell stage, which supplies an output-voltage signal under the control of said signal generated by the local oscillator. The Gilbert-cell stage further comprises an output load for carrying out a current-to-voltage conversion and supplying the output-voltage signal. Said transconductor stage further comprises a differential feedback network configured for reproducing said input-voltage signal on a differential load included in said voltage-to-current conversion module.

Description:
BACKGROUND OF THE INVENTION  
       [0001]     1. Field of the Invention  
         [0002]     The present description relates to apparatuses and methods for modulation of base-band signals into signals operating at given frequencies, in particular radio-frequency.  
         [0003]     2. Description of the Related Art  
         [0004]     The tendency of the market of microelectronic devices of presenting increasingly high levels of performance at extremely contained costs, as well as the concentration of an increasingly higher number of functions in portable devices, intensifies the demand for devices that involve a low power consumption. Said requirements are met via the use of high-efficiency CMOS technological processes with sub-micrometric channel lengths, which, however, on account of certain technological constraints, require very low supply voltages. All of these elements have in general a marked impact on the design of analog circuits, and frequently impose the use of non-conventional structures.  
         [0005]     One of the circuit blocks that is most affected by the new constraints of supply voltage is the transmission mixer (or modulator), which performs the function of converting to radio-frequency (RF) the low-frequency signal coming from the base-band (BB) circuit, with a conversion gain generally not lower than 0 dB. In addition, since the most modern modulation systems (CDMA, WLAN, etc.), characterized by encodings that generate instantaneously input signals of high amplitude, the use is necessary of circuit topologies with high input dynamic ranges.  
         [0006]     In what follows, known architectures for transmission modulators will be described, evaluating, for each of them, the minimum supply voltage that can be used, the corresponding power consumption, and the levels of performance.  
         [0007]      FIG. 1  represents a circuit diagram of a modulator apparatus, designated as a whole by the reference number  10 , representing the topology most widely used for the transmission modulator.  
         [0008]     As can be also appreciated in what follows, the modulator apparatus  10  basically comprises two branches corresponding to the two input nodes a and b applied on which is an input-voltage base-band differential signal V in  to be modulated. The modulator apparatus described here and the ones illustrated in what follows present a symmetrical architecture on said branches; hence, elements that are the same associated to each of said branches will be distinguished by the subscripts a and b.  
         [0009]     Said modulator apparatus  10  hence comprises a conversion module  20 , which, in this case, constitutes autonomously a transconductor stage. Said conversion module  20 , as has been mentioned, receives the input-voltage band-base signal V in  on the nodes a and b from an band-base operating apparatus (not shown in  FIG. 1 ). The nodes a and b correspond to the gate electrodes of respective conversion transistors M 1a  and M 1b , of a MOSFET type, equipped with respective degeneration resistances, R EEa  and R EEb , connected between their source electrode and ground. The drain electrodes of said conversion transistors M 1a  and M 1b  constitute the current outputs of the transconductor stage  20 .  
         [0010]     The modulator apparatus  10  represented in  FIG. 1  is a modulator of the so-called “stacked” type, in so far as the transconductor stage  20  shares the biasing current with a Gilbert-cell stage  30 , also referred to as “Gilbert Quad”. Said Gilbert cell  30  is of the double-balanced type and hence comprises a first pair of transistors M Qa  and M Qb  having their source electrode in common, as well as a second pair of transistors M Qc  and M Qd  connected in a similar way. A differential control signal V LO , produced by a local oscillator (not represented in  FIG. 1  either), is sent at input to the gate electrodes, associated in a common node, of the transistors M Qb  and M Qc . The drain electrodes of the transistors M Qa  and M Qd , according to the known circuit diagram of the balanced Gilbert cell, are connected to the drain electrodes, respectively, of the transistor M Qc  and of the transistor M Qb  and are also connected to the supply voltage V DD  via respective load resistances R La  and R Lb , which convert the current into voltage and across which the output-voltage signal V out  is then taken, whilst the Gilbert cell  30  is connected to the output of the transconductor stage  20  by means of the source electrodes of the first pair of transistors M Qa  and M Qb  and of the second pair of transistors M Qc  and M Qd , associated, respectively, to the drain electrode of the conversion transistors M 1a  and M 1b . Functionally the transconductor stage  20  carries out voltage-to-current conversion of the input-voltage signal V in  supplied by the base-band circuit, whilst the Gilbert cell  30 , stimulated by the control signal V LO  coming from the local oscillator thereof carries out the frequency conversion. The resulting RF current signal is then converted into the output-voltage signal V out  through the output load determined by the load resistances R La  and R Lb .  
         [0011]     From simple circuit considerations the following approximate expressions are obtained for a minimum supply voltage V DDmin  allowed and for a current consumption I SUPPLY  of the modulated apparatus  10  of  FIG. 1 :  
               V     DD   ⁢   min       =       V     i   ⁢           ⁢   n       +     V     DS   ⁢           ⁢   min       +       V   LO     2     +     V     DS   ⁢           ⁢   min       +         V     i   ⁢           ⁢   n       ·   G   ·   π     4     +         V     i   ⁢           ⁢   n       ·   G     2               (   1   )                 I   SUPPLY     =       π   ·     V     i   ⁢           ⁢   n       ·   G       2   ·     R   L                 (   2   )             
 
         [0012]     Clearly, in Eq. (1) of the input-voltage signal V in  the peak value is used, as likewise of the control signal V LO  coming from the local oscillator the amplitude is used. The reference V DSmin  designates a minimum value of the drain-to-source voltage at which the MOSFETs operate in the saturation region. The reference G designates the input-output voltage conversion gain of the modulator apparatus  10  (VN), and RL corresponds to the value of the load resistor R La,b .  
         [0013]     In order to simplify evaluation of the different known topologies of modulators and to compare them with the solution proposed, for each architecture the numeric value of the minimum supply voltage V DDmin  and of the current consumption I SUPPLY is calculated assuming the following set of circuit parameters: 
 
 G= 1(0 dB),  V   in (peak)=400 mV,  R   La,b   =100 Ω, V   Dsmin =200 mV,  V   LO (peak)=500 mV,  V   TH= 500 mV,  V   GS =600 mV.   (3) 
 
         [0014]     The above set (3) of circuit parameters is provided purely by way of example and as tool useful for performing a rapid comparison through a reasonable example of application. It does not constitute, hence, a limitation of the field of use of the invention.  
         [0015]     By substituting in Eqs. (1) and (2) the values of the set of parameters (3) it is obtained a minimum supply voltage V DDmin  of 1.56 V, a minimum current consumption I SUPPLY  of 8 mA, and a respective dissipated power of 9.8 mW.  
         [0016]     It should be noted how, even though the biasing current is shared between the transconductor stage  20  and the Gilbert cell  30 , which operates as mixer, the power consumption is relatively high. In addition, the minimum value of supply voltage V DDmin  that guarantees operation of said circuit topology is rather high, and this constitutes an even more limiting factor for modern sub-micrometric technologies.  
         [0017]      FIG. 2  shows a modulator apparatus  110  made according to another known architecture, the so-called “folded mixer” architecture. In a way similar to the apparatus  10  of  FIG. 1 , the modulator apparatus  110  comprises a transconductor stage  120  followed by the Gilbert cell  30 . However, the transconductor stage  120  comprises a voltage-to-current conversion module  20  set with the drain electrodes of the transistors M 1a  and M 1b  connected to current generators I a  and I b , which are in turn connected to the supply voltage V DD . In turn, the Gilbert cell  30  has its own load resistor R La  and R Lb  connected to the ground node. This enables improvement in the voltage dynamic range both of the input signal and of the output signal. The current generators  1   a  and  1   b  have the dual function of supplying the biasing current to both of the functional sub-blocks, i.e., the transconductor stage  20  and the Gilbert cell  30 , and of maximizing the signal transfer thereof thanks to their intrinsic high output impedance.  
         [0018]     The modulator apparatus  110  of  FIG. 2  enables very low supply voltages. In fact, the expressions of the minimum supply voltage V DDmin  allowed and of the corresponding current consumption I SUPPLY  are:  
               V     DD   ⁢           ⁢   min       =       V     DS   ⁢           ⁢   min       +       V   LO     2     +     V     DS   ⁢           ⁢   min       +         V     i   ⁢           ⁢   n       ·   G   ·   π     4     +         V     i   ⁢           ⁢   n       ·   G     2               (   4   )                 I   SUPPLY     =       π   ·     V     i   ⁢           ⁢   n       ·   G       R   L               (   5   )             
 
         [0019]     From a comparison of Eq. (4) with Eq. (1) and of Eq. (5) with Eq. (2) it emerges that, however advantageous the folded-mixer architecture may be in terms of dynamic range, and hence of minimum supply voltage allowed, it presents a current consumption that is double with respect to the architecture represented in  FIG. 1 .  
         [0020]     In fact, if the set of parameters (3) is inserted in Eqs. (4) and (5), it is obtained a minimum supply voltage V DDmin  of 1.16 V, but a minimum current consumption I SUPPLY  of 12.56 mA, which in practice corresponds to a dissipated power of 14.6 mW. It hence follows that, even though the circuit topology of  FIG. 2  is suitable for low-supply-voltage applications, it does not enable a very contained power dissipation, which is a fundamental parameter for portable applications.  
         [0021]      FIG. 3  represents a modulator apparatus  210  made according to a so-called “Gm-folded” architecture. A substantially similar modulator apparatus is also known from the document U.S. Pat. No. 5,172,079.  
         [0022]     In said circuit configuration a transconductor stage  220  comprises the usual voltage-to-current conversion module  20 , comprising the conversion transistors M 1a  and M 1b , in this case PMOS transistors, associated to corresponding degeneration resistor R EEa  and R EEa,b , set connected between their source electrodes and the supply voltage V DD . In addition, the transconductor stage  220  comprises in this case a first current mirror  225 , connected to the drain electrode of the conversion transistor M 1a  and made up of the transistors of M 2a ′ and M 2a ″, and a second current mirror  226 , connected to the source electrode of the transistor M 2a  and made up of the transistors M 2b ′ and M 2b ″. The current mirrors  225  and  226  mirror the current in the Gilbert cell  30 , that is, of the type similar to the one illustrated with reference to  FIG. 1 . The mirror factor N of the current mirrors  225  and  226 , defined as ratio between the aspect ratio of their transistors, is generally identified as a compromise between current consumption and output noise.  
         [0023]     The expressions of the minimum supply voltage V DDmin  and of the current consumption I SUPPLY  for the modulator apparatus  210  according to the Gm-folded topology are:  
               V     DD   ⁢           ⁢   min       =     MAX   ⁢     {             V     DS   ⁢           ⁢   min       +       V   LO     2     +     V     DS   ⁢           ⁢   min       +         V     i   ⁢           ⁢   n       ·   G   ·   π     4     +         V     i   ⁢           ⁢   n       ·   G     2                   V     i   ⁢           ⁢   n       +     V     DS   ⁢           ⁢   min       +     V   GS                         (   6   )                 I   SUPPLY     =         N   +   1     N     ·       π   ·     V     i   ⁢           ⁢   n       ·   G       2   ·     R   L                   (   7   )             
 
         [0024]     By substituting the set of parameters (3) in the above Eqs. (6) and (7) and setting the mirror ratio N to 5, we obtain a minimum supply voltage V DDmin  of 1.2 V and a current consumption I SUPPLY  of 7.54 mA, which in practice correspond to a dissipated power of 9 mW. Therefore, in this case, the circuit topology is satisfactory both from the standpoint of the minimum supply voltage allowed (even though it is not the minimum amongst the architectures proposed) and from the standpoint of the associated power dissipation. On the other hand, however, it presents a series of disadvantages that reduce the aforesaid advantages in the actual definition of the circuit. In fact, since the mirrors used in the transconductor are “simple”, i.e., not in “cascode” configuration, on account of the phenomenon of channel modulation, the transfer of signal, as likewise the replica of the biasing current, are markedly dependent upon the differences in the drain-to-source voltages between the transistors M 2a ′ and M 2a ″, as well as M 2b ′ and M 2b ″. In addition, on account of the rectification of the signal coming from the local oscillator on the source electrodes of the transistors of the Gilbert cell, the differences between said drain-to-source voltages of the transistors M 2a ′-M 2a ″ and M 2b ′-M 2b ″ are also a function both of the biasing voltage of the Gilbert cell  20  and of the amplitude of the control signal V LO  of the local oscillator. Said elements unfavorably affect the transfer of signal and determine a lack of accuracy of the conversion gain of the modulator apparatus; this represents in general a limit for completely integrated circuit applications. Furthermore, even though the effect of channel modulation can be mitigated by the use of long-channel MOS devices, since the constraints on the input and output dynamic ranges (low overdrive→high shape ratio W/L) impose the use of high shape factors, this results in large parasitic capacitances at the drains of the transistors M 2a ″ and M 2b ″, with consequent high feed-through of the local oscillator.  
       BRIEF SUMMARY OF THE INVENTION  
       [0025]     One embodiment of the present invention solves the drawbacks described above and proposes a solution that enables operation with a low value of minimum supply voltage and at low consumption levels, controlling way the gain in a precise, in particular in a way that presents low sensitivity in regard to the phenomenon of channel modulation.  
         [0026]     One embodiment of the present invention is an apparatus having the characteristics recalled in the claims, which form an integral part of the technical teachings regarding embodiments of the invention. One embodiments of the present invention regards also a corresponding method of modulation.  
     
    
     BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS  
       [0027]     Embodiments of the invention will now be described, purely by way of non-limiting example, with reference to the figures of the annexed plate of drawings, wherein:  
         [0028]      FIGS. 1, 2  and  3 , which regard known apparatuses, have already been described in the foregoing; and  
         [0029]      FIG. 4  shows an embodiment of a modulator apparatus according to one embodiment of the invention. 
     
    
     DETAILED DESCRIPTION OF THE INVENTION  
       [0030]     In brief, a modulator apparatus and a corresponding method of modulation are proposed, which envisage providing in the transconductor stage a feedback of a differential type for reproducing the input signal on the differential load, and a common-mode feedback so that the gain of the current mirror will be extremely accurate and will behave equivalently to a cascode mirror.  
         [0031]      FIG. 4  represents a circuit implementation of the modulator apparatus proposed, designated as a whole by the reference number  310 .  
         [0032]     Said modulator apparatus  310  comprises a transconductor stage  320 , which carries out the voltage-to-current conversion of the input-voltage signal V in  supplied by the base-band circuit. Associated to the output of the transconductor stage  320  is a Gilbert cell  30 , similar to the ones shown in  FIGS. 1 and 3 . Said Gilbert cell  30 , stimulated by the control signal V LO  coming from a local oscillator, which is not shown here either, carries out the conversion to the higher radio-frequency, or a frequency up-conversion, of the current signal, which is then transformed into an output-voltage signal V out  by means of the output load, which is represented schematically in  FIG. 4  by the resistances R La  and R Lb .  
         [0033]     As has been said, the transconductor stage  320  carries out the voltage-to-current conversion by means of the pairs of transistors M 1a -M 1b  and M 2a ′ and M 2b ′, and of the respective degeneration resistor R EEa  and R EEb  connected between the drain electrodes of the transistors M 1a  and M 1b  and the supply voltage V DD . However, in the transconductor stage  320 , a differential amplifier A 1  is connected to the input nodes a and b and hence receives at its input terminals the input-voltage signal V in . The outputs of said differential amplifier A 1  controls the gate terminals of the transistors M 2a ′ and M 2b ′, which, in turn, generate a differential current signal, which traverses the transistors M 1a  and M 1b , arranged in common-gate configuration, and finally generates, on the differential load constituted by the degeneration resistor R EEa  and R Eeb , a signal proportional to the input signal V IN . The transconductor stage  320  comprises in fact a differential-feedback network, which includes the conversion transistors M 1a  and M 1b  and the degeneration resistor R EEa,b , which are connected to the input signal V in  via resistance dividers R 1a -R 2a  and R 1b -R 2b  on the input terminals of the differential amplifier A 1 . The source electrodes of the conversion transistors M 1a  and M 1b  are moreover connected to the drain electrodes of the transistors M 2a ′ and M 2b ′. Departing from said drain electrodes are moreover respective first common-mode resistances R CMa ′ and R CMb ′, which connect up in a common-mode node CM. On said common-mode node CM a common-mode voltage V CM  is set up. Likewise, departing from the drain electrodes of the transistors M 2a ″ and M 2b ″ are respective second common-mode resistances R CMa ″ and R CMb ″, which connect up in a reference node REF. On said reference node REF a reference voltage V REF  is set up. The reference voltage V REF  and common-mode voltage V CM  constitute the inputs of a second differential amplifier A 2 , the output of which is connected to the gate electrodes of the common-gate transistors M 1a  and M 1b . Consequently, the first and second common-mode resistances, the transistors of the current mirrors  325  and  326 , as well as the second differential amplifier A 2  configure a common-mode feedback.  
         [0034]     If we assume for simplicity of exposition that, as regards the values of the resistances, we have R 2a =R 1a &gt;&gt;R EEa  and R 2b =R 1b &gt;&gt;R EEb , it may be noted from an examination of the transconductor stage  320  that the function of the differential feedback is to reproduce faithfully the input-voltage signal V IN  on the differential load constituted by the degeneration resistor R EEa  and R EEb  through direct control of the differential voltage applied to the gates of the transistors M 2a ′, M 2b ′. In addition, as a result of the resistances R EEa,b , the differential current supplied by the transistors M 1a , M 1b  is directly proportional to the differential-voltage signal across the resistances themselves. Said differential-current signal is then mirrored, with a mirror ratio N:1, defined by the ratio between the shape factors of the transistors M 2a,b ′-M 2a,b ″, on the circuit branches which arrive at the Gilbert cell  30 . The presence of the common-mode feedback, performed by the operational amplifier A 2 , ensures that the current mirror is extremely accurate and will behave equivalently to a cascode mirror. In fact, the common-mode feedback controls the biasing voltage of the gate of the transistors M 1a  and M 1b  so as to nullify the difference between the reference voltage V REF  and common-mode voltage V CM  at its input. The advantage that derives therefrom is that, irrespective of the signal-rectification effect on the source of the transistors of the Gilbert cell  30 , the mean value of the differences between the drain-to-source voltages of the transistors M 2a ′-M 2a ″ and M 2b ′-M 2b ″ is zero, and, consequently, the modulator apparatus  310  is not affected by the problems of accuracy of conversion gain. Furthermore, said improvement in the accuracy of the current mirror introduced by the common-mode feedback makes it possible to use small channel lengths for the transistors of one and the same mirror, to the advantage of the dynamic range, the speed of the transconductor stage, and, seeing that the overall capacity on the drain electrodes of the transistors M 2a ″ and M 2b ″ of the second current mirror  326  can be consequently rendered negligible, also to the advantage of the feed-through of the local oscillator.  
         [0035]     The expressions of the minimum supply voltage V DD  and of the current consumption I SUPPLY  of the invention proposed are the following:  
               V     DD   ⁢               =     MAX   ⁢     {             V     DS   ⁢           ⁢   min       +       V   LO     2     +     V     DS   ⁢           ⁢   min       +         V     i   ⁢           ⁢   n       ·   G   ·   π     4     +         V     i   ⁢           ⁢   n       ·   G     2                   V     i   ⁢           ⁢   n       +     V     DS   ⁢           ⁢   min       +     V     DS   ⁢           ⁢   min                           (   8   )                 I   SUPPLY     =         N   +   1     N     ·       π   ·     V     i   ⁢           ⁢   n       ·   G       2   ·     R   L                   (   9   )             
 
         [0036]     By substituting the set of parameters (3) in Eqs. (8) and (9), and setting the mirror ratio N to 5, it is obtained a minimum supply voltage V DD  of 1.16 V and a current consumption of 7.54 mA, to which there in practice corresponds a dissipated power of 8.75 mW.  
         [0037]     Hence, the proposed embodiment is an improvement compared to the ones examined in the known art both as regards the minimum supply voltage and as regards power consumption, and moreover presents multiple advantages that reflect positively on the feed-through and on the accuracy of the conversion gain.  
         [0038]     Consequently, without prejudice to the principle of the invention, the details of construction and the embodiments may vary, even significantly, with respect to what is described and illustrated herein, purely by way of non-limiting example, without thereby departing from the scope of the invention, as defined in the ensuing claims.  
         [0039]     For example, with respect to the embodiment described, the resistances can be replaced by generic impedances, and the output load, which is resistive, can be replaced by a generic load, e.g., an active load, an inductive passive load, etc.  
         [0040]     The architecture proposed can be used satisfactorily both for providing individual modulators and for providing I/Q modulators.  
         [0041]     In addition, one embodiment can be applied also as down-converter in the receiver chain of modules in certain dual-conversion applications, provided that the input intermediate frequency (IF) is sufficiently low as compared to the band-gain product of the differential amplifier.  
         [0042]     Even though the circuit described above has been developed using only NMOS transistors, the invention can be extended also to the use of bipolar transistors, with BiCMOS approach, and of dual architectures (PMOS transistors, pnp transistors).  
         [0043]     All of the above U.S. patents, U.S. patent application publications, U.S. patent applications, foreign patents, foreign patent applications and non-patent publications referred to in this specification and/or listed in the Application Data Sheet, are incorporated herein by reference, in their entirety.