Abstract:
A drive state detection circuit is disclosed that detects a drive state of plural parts driven by alternating currents. The drive state detection circuit comprises current detecting parts to detect alternating current detection signals of the respective alternating currents flowing through the driven parts, a maximum value output part to output an alternating current detection signal having a maximum value among the alternating current detection signals detected by the current detecting parts, a coefficient multiplication part to multiply the signal output from the maximum value output part by a coefficient, a comparing part to compare the multiplied signal with the alternating current detection signals so as to output state signals corresponding to the respective driven parts, a logic synthesizing part to generate an output by logically synthesizing the state signals output from the comparing part, and an output part to generate a drive state detection signal based on the output from the logic synthesizing part.

Description:
BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The present invention relates to a drive state detection circuit, and particularly relates to a drive state detection circuit that detects a drive state of parts driven by alternating currents. 
   2. Description of the Related Art 
   CCFLs (Cold Cathode Fluorescent Lamps) are used as, for example, back lights of liquid crystal display monitors. The CCFLs are driven by a drive system with an alternating current. The drive system is equipped with a protection system that detects a drive state of the CCFLs to protect the CCFLs. 
   The protection system usually detects the drive state of CCFLs by outputting maximum values of the voltage and current supplied to the CCFLs. For outputting maximum values, a protection system disclosed, for example, in Japanese Patent Laid-Open publication No. 6-267674 and No. 2002-134293, holds AC signals of the voltage and current supplied to CCFLs at their peaks, and converts them into DC signals. 
   State detection circuits for CCFLs with the above type of protection systems therefore need to have additional circuits such as peak hold circuits, which make the configuration complicated. 
   SUMMARY OF THE INVENTION 
   A general object of the present invention is to provide a drive state detection circuit to overcome at least one disadvantage described above. A specific object of the present invention is to provide a simply-configured drive state detection circuit capable of detecting a drive state of driven parts without having additional circuits. 
   According to an aspect of the present invention, there is provided a drive state detection circuit to detect a drive state of plural parts driven by alternating currents, comprising current detecting parts to detect alternating current detection signals of the respective alternating currents flowing through the driven parts, a maximum value output part to output an alternating current detection signal having a maximum value among the alternating current detection signals detected by the current detecting parts, a coefficient multiplication part to multiply the signal output from the maximum value output part by a coefficient, a comparing part to compare the multiplied signal with the alternating current detection signals so as to output state signals corresponding to the respective driven parts, a logic synthesizing part to generate an output by logically synthesizing the state signals output from the comparing part, and an output part to generate a drive state detection signal based on the output from the logic synthesizing part. 
   It is preferable that the drive state detection circuit further comprise a reference voltage generating part to generate a reference voltage corresponding to a lower limit of a maximum value, and a second comparing part to compare the signal output from the maximum value output part with the reference voltage generated by the reference voltage generating part and output a state signal based on a comparison result to the logic synthesizing part. 
   It is also preferable that the drive state detection circuit further comprise a voltage detecting part to detect voltages applied to the driven parts, a second reference voltage generating part to generate a reference voltage corresponding to an upper limit of a voltage applied to the driven parts, and a third comparing part to compare the voltages detected by the voltage detecting parts with the reference voltage generated by the second reference voltage generating part so as to output a state signal based on a comparison result to the logic synthesizing part. 
   It is also preferable that the output part be adapted to invert the drive state detection signal when the output from the logic synthesizing part stays in a predetermined condition during a predetermined time. 
   It is also preferable that the output part comprise a capacitor, a charging part to charge the capacitor, a discharging part to discharge the capacitor according to the output of the logic synthesizing part, and an output circuit to invert the drive state detection signal according to a charging voltage of the capacitor. 
   It is also preferable that the plural driven parts be Cold Cathode Fluorescent Lamps. 
   According to the present invention, the highest level signal among input signals can be selectively output by utilizing switching characteristics of plural input transistors, a current mirror circuit, and an output transistor. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a circuit diagram of a CCFL lighting system according to an embodiment of the present invention; 
       FIG. 2  is a circuit diagram of a driver IC; 
       FIG. 3  is a circuit diagram of a PWM control section; 
       FIG. 4  is a circuit diagram of a protection circuit section; 
       FIG. 5  is an operations chart of the protection circuit section; 
       FIG. 6  is another operations chart of the protection circuit section; 
       FIG. 7  is a circuit diagram of a maximum value output circuit; and 
       FIG. 8  is an operations chart of the maximum value output circuit. 
   

   DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
   This embodiment describes a case where a maximum value output circuit is used in a CCFL (Cold Cathode Fluorescent Lamp) lighting system. Firstly, the CCFL lighting system is described below. 
   [System Configuration] 
     FIG. 1  is a circuit diagram of a CCFL lighting system  1  according to this embodiment of the present invention. 
   The CCFL lighting system  1  of this embodiment is used as, for example, a back light system of a liquid crystal display monitor, comprising a CCFL section  11 , a resonant circuit  12 , a driver IC (Integrated Circuit)  13 , a protection IC (Integrated Circuit)  14 , a peak hold circuit  15 , a reference voltage generator  16 , and capacitors C 1  and C 2 . 
   The CCFL section  11  comprises paired CCFLs  21  and  22  arranged in parallel. The paired CCFL  21  includes two CCFLs  31  and  32  arranged in parallel. The paired CCFL  22  also includes two CCFLs  41  and  42  arranged in parallel. 
   An end of each of the CCFLs  31 ,  32 ,  41  and  42  is connected to the resonant circuit  12 . The other end of each of the CCFLs  31  and  32  is grounded through detection resistances Rs 1  and Rs 2 . The other end of each of the CCFLs  41  and  42  is also grounded through detection resistances Rs 3  and Rs 4 . 
   When a voltage with a predetermined frequency (e.g. 50 kHz) is applied to each end of the CCFLs  31 ,  32 ,  41  and  42 , drive currents flow therethrough and the CCFLs  31 ,  32 ,  41  and  42  are turned on. When a voltage with a frequency higher (e.g. 100 kHz) or lower than the predetermined frequency is applied, the CCFLs  31 ,  32 ,  41  and  42  are turned off. 
   The resonant circuit  12  receives a drive signal with a predetermined frequency from the driver IC  13 . The resonant circuit  12  includes a capacitor and a transformer (not shown). With the capacitance and inductance thereof, the resonant circuit  12  resonates with the drive signal from the driver IC  13 , and provides drive power to the CCFL section  11 . 
   [Driver IC  13 ] 
     FIG. 2  is a circuit diagram of the driver IC  13 . 
   The driver IC  13  comprises a VCO (Voltage Control Oscillator) circuit  51 , a starting circuit  52 , an error amplifier  53  and a voltage control circuit  54 . 
   A control terminal Tcnt of the VCO circuit  51  is connected to the starting circuit  52 , the error amplifier  53 , the voltage control circuit  54  and a terminal T 4 . The VCO circuit  51  outputs oscillation with a frequency corresponding to the voltage applied to the control terminal Tcnt from an output terminal Tosc. 
   The output terminal Tosc of the VCO circuit  51  is connected to an output terminal T 1  of the driver IC  13 . The oscillation of the VCO circuit  51  is output from the output terminal T 1  to the resonant circuit  12 . 
   The starting circuit  52  controls the control voltage of the VCO circuit  51  so as to quickly turn on the CCFLs  31 ,  32 ,  41  and  42  at the time of, for example, power-on. 
   The error amplifier  53  has an inverting input terminal connected to a terminal T 2  and a non-inverting input terminal connected to a terminal T 3 . The terminal T 2  receives an average value signal from the protection IC  14 , while the terminal T 3  receives a reference voltage from a reference voltage generator. The error amplifier  53  outputs a voltage corresponding to a difference between the average value signal and the reference voltage. The output of the error amplifier  53  is sent to the control terminal Tcnt of the VCO circuit  51  and the terminal T 4 . 
   The voltage control circuit  54  is connected to a terminal T 5 . The terminal T 5  is connected to a terminal T 14  of the protection IC  14 , from which a stop signal is provided. The voltage control circuit  54  keeps the control terminal Tcnt of the VCO circuit  51  at a high level according to the stop signal from the protection IC  14 . Once the voltage control circuit  54  keeps its output high level, the output remains high level until it is reset by a power-off. 
   The terminal T 4  is connected to a terminal T 15  of the protection IC  14  and an end of the capacitor C 1 . The control voltage applied to the control terminal Tcnt is controlled by a charging voltage of the capacitor C 1 , so that the oscillation frequency of the VCO circuit  51  is controlled. 
   [Protection IC  14 ] 
   Referring to  FIG. 1 , the protection IC  14  comprises a PWM (Pulse Width Modulation) control section  61  and a protection circuit section  62 . The PWM control section  61  performs PWM control on the oscillation state of the VCO circuit  51  of the driver IC  13  according to the luminance of the CCFLs  31 ,  32 ,  41  and  42 . 
   [PWM Control Section  61 ] 
     FIG. 3  is a circuit diagram of the PWM control section  61 . 
   The PWM control section  61  comprises a triangular wave generating circuit  71 , a comparator  72 , a gate circuit  73 , analog switch  74 , a discharge switch  75 , a comparator  76 , resistances R 11 , R 12  and R 13 , and a capacitor C 11 . 
   A terminal T 17  receives a specified luminance signal for determining a luminance from an external unit. The specified luminance signal received by the terminal T 17  is provided to an inverting input terminal of the comparator  72 . A non-inverting input terminal of the comparator  72  receives a triangular wave from the triangular wave generating circuit  71 . The comparator  72  compares the luminance signal with the triangular wave. If the triangular wave is higher than the luminance signal, the output of the comparator  72  becomes high level. If the triangular wave is lower than the luminance signal, the output becomes low level. The comparator  72  thus generates a pulse corresponding to the frequency of the triangular wave and having a pulse width corresponding to the luminance signal. 
   The output pulse of the comparator  72  is provided to the switch  75  through a delay circuit including the resistance R 11  and the capacitor C 11 , and also provided to the gate circuit  73 . The switch  75 , which is arranged between the terminal T 15  and the ground, is turned on/off with a pulse delayed from the output pulse of the comparator  72  by a time determined by the resistance R 11  and the capacitor C 11 . When the pulse is low level, the switch  75  is switched off and charges the capacitor. When the pulse is high level, the switch  75  is switched on and discharges the capacitor C 1 . 
   The gate circuit  73  inverts the output pulse of the comparator  72  to input therein. The gate circuit  73  also receives the output of the comparator  76 . The gate circuit  73  outputs a logical AND of the inverted output of the comparator  72  with the output of the comparator  76 . The output of the gate circuit  73  is provided to the analog switch  74 . 
   The analog switch  74 , which is arranged between the terminals T 15  and T 16 , is turned on/off according to the output of the gate circuit  73 . When the output of the gate circuit  73  is high level, the analog switch  74  is turned on to short-circuit the terminals T 15  and T 16 . When the output of the gate circuit  73  is low level, the analog switch  74  is turned off to open-circuit the terminals T 15  and T 16 . 
   An inverting input terminal of the comparator  76  is connected to the terminal T 16 , while a non-inverting input terminal of the comparator  76  is connected to a connection point between the resistances R 12  and R 13 . An end of the resistance R 12  is connected to the non-inverting input terminal of the comparator  76 , while the other end is connected to the terminal T 15 . An end of the resistance R 13  is connected to a connection point between the non-inverting input terminal of the comparator  76  and the end of the resistance R 12 , while the other end of the resistance R 13  receives a supply voltage Vdd. 
   The comparator  76  compares a potential of the connection point between the resistances R 12  and R 13  with a potential of the terminal T 16 . If the potential of the connection point between the resistances R 12  and R 13  is higher than the potential of the terminal T 16 , the output becomes high level. If the potential of the connection point between the resistances R 12  and R 13  is lower than the potential of the terminal T 16 , the output becomes low level. 
   The following describes the operations of the PWM control section  61 . 
   First, when the output pulse of the comparator  72  becomes high level, the switch  74  is turned off to open-circuit the terminals T 15  and T 16 . The capacitor C 1  is therefore electrically disconnected from the capacitor C 2 . 
   Then, the switch  75  becomes high level with a delay caused by the resistance R 11  and capacitor C 11 . The switch  75  is therefore turned on shortly after the output of the comparator  72  becomes high level. As the switch  75  is turned on, the capacitor C 1  connected to the terminal T 15  is discharged. 
   When the capacitor C 1  is discharged, the potential of the terminal T 15  is lowered. Then, the triangular wave output from the triangular wave generating circuit  71  is lowered. The output of the comparator  72  therefore becomes low level, and the switch  75  is turned off with a little delay caused by the resistance R 11  and the capacitor C 11 . As the switch  75  is turned off, the capacitor C 1  is charged by the potential of the terminal T 4  of the driver IC  13 . 
   When the capacitor C 1  is charged, the potential of the terminal T 15  is raised. As the potential of the terminal T 15  is raised, the potential of the non-inverting input terminal of the comparator  76  is raised. 
   When the potential of the non-inverting input terminal of the comparator  76  exceeds the potential of the terminal T 16 , i.e., the charging voltage of the capacitor C 2 , the output of the comparator  76  becomes high level. Then, the output of the gate circuit  73  becomes high level, and the analog switch  74  is turned on. As the switch  74  is turned on, the capacitor C 1  is electrically connected to the capacitor C 2 . The terminal T 4  is therefore electrically connected to the capacitors C 1  and C 2 . 
   As such, when the charging voltage of the capacitor C 1  reaches a desired voltage with respect to the charge voltage of the capacitor C 2 , the analog switch  74  is turned on to electrically connect the terminal T 4  to the capacitors C 1  and C 2 . This can prevent an overshoot of the capacitor C 1  during charging. 
   When the triangular wave of the triangular wave generating circuit  71  becomes higher than the specified luminance signal, the output of the comparator  72  becomes high level. Then, the output of the gate circuit  73  becomes low level, and the analog switch  74  is turned off. As the analog switch  74  is turned off, the capacitor C 2  retains the potential of the terminal T 4  of the driver IC  13 . Shortly after the analog switch  74  is turned off, the switch  75  is turned on to discharge the capacitor C 1 . Since the analog switch  74  is already turned off, the capacitor C 2  keeps retaining the potential of the terminal T 4 . 
   With these operations, the potential of the terminal T 4  of the driver IC  13  is controlled so as to vary forming pulses according to the output pulse of the comparator  72 . 
   Since the potential of the terminal T 4  varies forming pulses, the driver IC  13  can change the oscillation frequency of the VCO circuit  51  so that the frequency is switched between approximately 50 kHz and approximately 100 kHz. When the output oscillation frequency of the VCO circuit  51  becomes 50 kHz, the resonant circuit  12  resonates therewith to turn on the CCFLs  31 ,  32 ,  41  and  42 . When the output oscillation frequency of the VCO circuit  51  becomes 100 kHz, the power supply from the resonant circuit  12  to the CCFLs  31 ,  32 ,  41  and  42  is stopped and the CCFLs  31 ,  32 ,  41  and  42  are turned off. 
   In this way, the power is intermittently supplied to the CCFLs  31 ,  32 ,  41  and  42 , so that the luminance is kept constant. 
   The overshoot of the potential of the terminal T 4  can be prevented by controlling the connection of the capacitors C 1  and C 2  and changing the capacity thereof by means of the analog switch  74  during the charging of the capacitor C 1 . Accordingly, the oscillation output of the VCO circuit  51 , of which oscillation frequency is controlled by the potential of the terminal T 4 , can be stabilized. 
   [Protection Circuit Section  62 ] 
   The following describes the protection circuit section  62 , which characterizes the present invention. 
   The protection circuit  62  is configured to detect a maximum value of a voltage and current supplied to the CCFL section  11 , and detect malfunction of the CCFL section  11 . 
     FIG. 4  is a circuit diagram of the protection circuit section  62 . 
   The protection circuit  62  comprises a maximum value output circuit  91 , a comparator  92 , a reference voltage generator  93 , a coefficient multiplication circuit  94 , comparators  95 ,  96  and  97 , a reference voltage generator  98 , an AND gate  99 , an output circuit  100 , and diodes D 1  and D 2 . 
   The maximum value output circuit  91  receives detection voltages through terminals T 12  and T 13 . The terminal T 12  is grounded through the diode D 1  connected in an opposite direction. The terminal T 13  is grounded through the diode D 2  connected in an opposite direction. 
   The diodes D 1  and D 2  function as protection elements for the protection IC  14 . The diodes D 1  and D 2  perform half-wave rectification of the detection voltages from the terminals T 12  and T 13 . The detection voltages are then provided to the maximum value output circuit  91 . 
   The maximum value output circuit  91  selectively outputs a higher one of the detection voltages provided from the terminals T 12  and T 13 . 
   The maximum value signal output from the maximum value output circuit  91  is provided to a non-inverting input terminal of the comparator  92  and the coefficient multiplication circuit  94 . An inverting input terminal of the comparator  92  receives a reference voltage from the reference voltage generator  93 . The reference voltage generated by the reference voltage generator  93  is set to a lower limit value of the maximum value signal. 
   If the maximum value signal from the maximum value output circuit  91  is greater than the reference voltage generated by the reference voltage generator  93 , the output of the comparator  92  becomes high level. If the maximum value signal is smaller than the reference voltage generated by the reference voltage generator  93 , the output of the comparator  92  becomes low level. The output of the comparator  92  is provided to the AND gate  99 . 
   Meanwhile, the coefficient multiplication circuit  94  multiplies the maximum value signal by 0.8. In other words, the coefficient multiplication circuit  94  outputs a signal equivalent to 80 percent of the maximum value. The signal multiplied by 0.8 by the coefficient multiplication circuit  94  is provided to inverting input terminals of the comparators  95  and  96 . 
   A non-inverting input terminal of the comparator  95  is provided with a detection signal V 12  from the terminal T 12 . If the detection signal V 12  is greater than the signal equivalent to 80 percent of the maximum value from the coefficient multiplication circuit  94 , the output of the comparator  95  becomes high level. If the detection signal V 12  is smaller, the output of the comparator  95  becomes low level. 
   A non-inverting input terminal of the comparator  96  is provided with a detection signal V 13  from the terminal T 13 . If the detection signal V 13  is greater than the signal equivalent to 80 percent of the maximum value from the coefficient multiplication circuit  94 , the output of the comparator  96  becomes high level. If detection signal V 13  is smaller, the output of the comparator  96  becomes low level. The outputs of the comparators  95  and  96  are provided to the AND gate  99 . 
   An inverting input terminal of the comparator  97  is provided with an output of the hold circuit  15  through a terminal T 11 . The hold circuit  15  holds the maximum voltage of a connection point between detection resistances Rs 1  and Rs 2  and a connection point between detection resistances Rs 3  and Rs 4 . A non-inverting input terminal of the comparator  97  receives a reference voltage from the reference voltage generator  98 . The reference voltage generated by the reference voltage generator  98  is set to a voltage corresponding to a maximum drive voltage. 
   If the output voltage of the hold circuit  15  is higher than the reference voltage from the reference voltage generator  98 , the output of the comparator  97  becomes low level. If the output voltage of the hold circuit  15  is lower than the reference voltage from the reference voltage generator  98 , the output of the comparator  97  becomes high level. The output of the comparator  97  is provided to the AND gate  99 . 
   The AND gate  99  is provided with the outputs of the comparators  92 ,  95 ,  96  and  97 . The AND gate  99  outputs a logical AND of the outputs of the comparators  92 ,  95 ,  96  and  97 . If all of the outputs of the comparators  92 ,  95 ,  96  and  97  are high level, the output of the AND gate  99  becomes high level. If any of the outputs of the comparators  92 ,  95 ,  96  and  97  is low level, the output of the AND gate  99  becomes low level. The output of the AND gate  99  is provided to the output circuit  100 . 
   The output circuit  100  comprises a current source  111 , a comparator  112 , a reference voltage generator  113 , a capacitor C 21 , and transistors M 11  and M 12 . 
   The output of the AND gate  99  is provided to a gate of the transistor M 11 . The transistor M 11  is an N-channel MOS field-effect transistor, having a grounded source. The capacitor C 21  is connected to the transistor M 11  in parallel with the drain-source thereof. The current source  111  supplies a charging current to a connection point between the drain of the transistor M 11  and the capacitor C 21 . 
   The transistor M 11  is turned on when the output of the AND gate  99  is high level, and is turned off when the output of the AND gate is low level. When the transistor M 11  is turned off, the capacitor C 21  is charged with the charging current from the current source  111 . When the transistor M 11  is turned on, the charge in the capacitor C 21  is released to the ground through the transistor M 11 . In this way, the capacitor c 21  is charged or discharged according to the on/off state of the transistor M 11 . 
   The charging voltage of the capacitor C 21  is applied to an inverting input terminal of the comparator  112 . A non-inverting input terminal of the comparator  112  receives a reference voltage from the reference voltage generator  113 . If the charging voltage of the capacitor C 21  is higher than the reference voltage from the reference voltage generator  113 , the output of the comparator  112  becomes low level. If the charging voltage of the capacitor C 21  is lower than the reference voltage from the reference voltage generator  113 , the output of the comparator  112  becomes high level. The output of the comparator  112  is provided to a gate of the transistor M 12 . 
   The transistor M 12  is an N-channel MOS field-effect transistor, having a source connected to the ground and a drain connected to the terminal T 14 . The transistor M 12  is turned on when the output of the comparator  112  is high level, and is turned off when the output is low level. 
   [Operations of Protection Circuit] 
     FIGS. 5 and 6  are operations diagrams of the protection circuit section  62 . In  FIG. 5 , (A) shows operations of the comparator  92 ; (B) shows operations of the comparators  95  and  96 ; and (C) shows operations of the comparator  97 . In  FIG. 6 , (A) shows the output of the comparator  92 ; (B) shows the output of the comparator  95  or  96 ; (C) shows the output of the comparator  97 ; (D) shows the output of the AND gate  99 ; (E) shows the charging voltage of the capacitor C 21 ; (F) shows the output of the comparator  112 ; and (G) shows the output of the terminal T 14 . A period T 10  in  FIG. 6  shows a normal operation state during normal operations, and a period T 20  shows a state when malfunction is detected. 
   [Normal Operation State] 
   Referring to (A) in  FIG. 5 , in a normal operation state, as shown an output voltage Vmax (continuous line) of the maximum value output circuit  91  exceeds a reference voltage Vref 1  generated by the reference voltage generator  93 . Therefore, the output of the comparator  92  becomes high level. Since the output of the comparator  92  is high level only when the output of the maximum value output circuit  91  is higher than the reference voltage Vref generated by the reference voltage generator  93 , the pulse shows an intermittent waveform as shown in (A) in  FIG. 6 . 
   Referring to (B) in  FIG. 5 , the comparators  95  and  96  compare a voltage V0.8 (dashed line) of 80 percent of the output voltage Vmax of the maximum value output circuit  91  with output voltages V 12  and V 13  of the terminal T 12  and T 13 , and output the comparison result. In the normal state, the output voltages V 12  and V 13  (chain double-dashed line) of the terminals T 12  and T 13  are higher than the voltage V0.8. The output voltages V 12  and V 13  of the terminals T 12  and T 13  and the voltage V0.8 show a half-wave rectification waveform, where an unstable period appears in approximately during a half of each cycle. If the output during the unstable period is recognized as low level, the pulse shows an intermittent waveform as shown in (B) in  FIG. 6 . 
   Referring to (C) in  FIG. 5 , the comparator  97  compares a DC voltage Vh from the hold circuit  15  with a reference voltage Vref 2  generated by the reference voltage generator  98 . In the normal operation state, the voltage Vh (chain dashed line) is lower than the reference voltage Vref 2 . Therefore, the output of the comparator  97  is high level. 
   As can be seen, in the normal operation state, during the period T 11  ((A) in  FIG. 6 ) in which the output of the comparator  92  is high level, all of the outputs of the comparators  92 ,  95 ,  96  and  97  are high level ((A), (B), (C) in  FIG. 6 ) The output of the AND gate  99  is therefore high level during the period T 11  as shown in (D) in  FIG. 6 . 
   When the output of the AND gate  99  becomes high level in the period T 11 , the transistor M 11  is turned on to discharge the charging voltage of the capacitor C 21  as shown in (E) in  FIG. 6 . In the period except the period T 11 , the output of the AND gate  99  becomes low level, so that the transistor M 11  is turned off to permit the current source  111  to charge the capacitor C 21 . Therefore, charging is performed in the intervals between the periods T 11  as shown in (E) in  FIG. 6 . 
   The charging voltage of the capacitor C 21  does not reach the reference voltage generated by the reference voltage generator  113 , because the period T 11  appears each cycle of the output voltages of the terminals T 12  and T 13 . As the charging voltage of the capacitor C 21  stays lower than the reference voltage Vref 3 , the output of the comparator  112  stays high level as shown in (F) in  FIG. 6 . The terminal T 14  therefore stays low level as shown in (G) in  FIG. 6 . 
   [State When Malfunction is Detected] 
   If the output of the maximum value output circuit  91  falls below the reference voltage, i.e., the lower limit value of the maximum value signal, the output of the comparator  92  becomes low level. 
   Referring to  FIG. 6 , in a period T 20 , when the output of the comparator  92  becomes low level as shown in (D), the output (continuous line) of the AND gate  99  becomes low level. While the output of the AND gate  99  stays low level and therefore the transistor M 11  is kept at the off state, the capacitor C 21  is charged by the current source  111  as shown in (E). When the charging voltage of the capacitor C 21  reaches the reference voltage Vref 3  at a time t 20 , the output of the comparator  112  becomes low level as shown in (F) and the terminal T 14  becomes high level as shown in (G). 
   If the voltage of the terminal T 12  or T 13  falls below the output of the coefficient multiplication circuit  94 , i.e., 80 percent of the maximum value signal due to malfunction related to the connection or light-out of the CCFL section  11 , the output of the comparator  95  or  96  (chain dashed line in (B)) stays low level during T 20 . 
   When the output of the comparator  92  becomes low level, the output of the AND gate  99  (continuous line in (D)) becomes low level in the period T 20 . While the output of the AND gate  99  stays low level and therefore the transistor M 11  is kept at the off state, the capacitor C 21  is charged by the current source  111  as shown in (E). When the charging voltage of the capacitor C 21  reaches the reference voltage Vref 3  at the time t 20 , the output of the comparator  112  becomes low level as shown in (F) and the terminal T 14  becomes high level as shown in (G). 
   If a voltage is excessively applied to the CCFL section  11  and the voltage of the terminal T 11  exceeds the reference voltage generated by the reference voltage generator  98 , the output of the comparator  97  (chain double-dashed line in (C)) becomes low level. When the output of the comparator  97  becomes low level, the output of the AND gate  99  (continuous line in (D)) becomes low level in the period T 20 . While the output of the AND gate  99  stays low level and therefore the transistor M 11  is kept at the off state, the capacitor C 21  is charged by the current source  111  as shown in (E). When the charging voltage of the capacitor C 21  reaches the reference voltage Vref 3  at the time t 20 , the output of the comparator  112  becomes low level as shown in (F) and the terminal T 14  becomes high level as shown in (G). 
   When the terminal T 14  becomes high level, the malfunction is detected by the driver IC  13 , which has the terminal T 5  connected to the terminal T 14 . When the terminal T 5  becomes high level, the voltage control circuit  54  stops the oscillation output of the VCO circuit  51 . 
   An average value circuit  101  is provided with the detection signals V 12  and V 13  from the terminals T 12  and T 13 . The average value circuit  101  generates a signal corresponding to the detection signals V 12  and V 13  and outputs the signal from the terminal T 18 . The terminal T 18  is connected to the terminal T 2  of the driver IC  13 . 
   [Maximum Value Output Circuit  91 ] 
   The following describes the maximum value output circuit  91 . 
     FIG. 7  is a circuit diagram of the maximum value output circuit  91 . 
   The maximum value output circuit  91  selectively outputs the maximum level signal among the detection signals provided from the terminals T 12  or T 13 , comprising bipolar transistors Q 11 , Q 12  and Q 13 , MOS field-effect transistors M 21  and M 22 , and a current source  121 . 
   The bipolar transistors Q 11 , Q 12  and Q 13  are NPN transistors. The transistors Q 11  and Q 12  have substantially the same characteristics. 
   The transistor Q 11 , serving as an input transistor, has a base connected to the terminal T 12 , a collector connected to a drain and gate of the transistor M 21  and to a gate of the transistor M 22 , and an emitter grounded through the current source  121 . The transistor Q 11  introduces a current corresponding to the detection signal of the terminal T 12  from the collector. 
   The transistor Q 12 , also serving as an input transistor, has a base connected to the terminal T 13 , a collector connected to the drain and gate of the transistor M 21  and to the gate of the transistor M 22 . The transistor Q 12  introduces a current corresponding to the detection signal of the terminal T 13  from the collector. 
   The transistor Q 13 , serving as an output transistor, has a collector connected to a drain of the transistor M 22 , and an emitter grounded through the current source  121 . The current flowing through the transistor Q 13  corresponds to the current flowing through the transistor Q 11  or Q 12 . 
   The transistors M 21  and M 22  are P-channel MOS field-effect transistors. The transistor M 21  has a source to which the supply voltage Vdd is applied, and a gate connected to a drain thereof and the gate of the transistor M 22 . The transistor M 22  has a source to which the supply voltage Vdd is applied, and a gate connected to the gate and drain of the transistor M 21 . The transistors M 21  and M 22  form a current mirror circuit so that a current corresponding to the current introduced from the collector of the transistor Q 11  or Q 12  is output from the drain of the transistor M 22 . The drain of the transistor M 22  is connected to the collector and base of the output transistor Q 13 . 
   The emitter of the transistor Q 13  is grounded through the current source  121 . The drain of the transistor M 22  and a connection point between the collector and base of the transistor Q 13  are maximum value outputs. The output of the maximum value output circuit  91  is provided to the comparator  92  and the coefficient multiplication circuit  94 . 
     FIG. 8  is an operations chart of the maximum value output circuit  91 . In  FIG. 8 , (A) shows an input signal, and (B) shows an output signal. 
   As shown in a period T 11  in (A), when the detection signal V 12  from the terminal T 12  is higher than the detection signal V 13  from the terminal T 13 , the collector current of the transistor Q 11  is well over the collector current of the transistor Q 12  due to the characteristics of the collector current with respect to the base-emitter voltage of the bipolar transistor. Namely, the current output capacity of the transistor Q 11  increases well over the current output capacity of the transistor Q 12 . 
   The transistors Q 11  and Q 12  are configured to provide a current to the same source, i.e., the current source  121 . Therefore, when the current output capacity of the transistor Q 11  is well over the current output capacity of the transistor Q 12 , most of the current provided to the current source  121  is from the transistor Q 11 . 
   As a current mirror circuit is formed by the transistors M 11  and M 12 , a current same as the current of the transistor Q 11  flows through the collector of the transistor Q 13 . Accordingly, a signal same as the signal appearing at the base of the transistor Q 11  appears at the base of the transistor Q 13  as shown in (B). 
   On the other hand, as shown in the period T 12  in (A), when the detection signal V 13  from the terminal T 13  is higher than the detection signal V 12  from the terminal T 12 , the collector current of the transistor Q 12  is well over the collector current of the transistor Q 11  due to the characteristics of the collector current with respect to the base-emitter voltage of the bipolar transistor. Namely, the current output capacity of the transistor Q 12  increases well over the current output capacity of the transistor Q 11 . 
   The transistors Q 11  and Q 12  are configured to provide a current to the same source, i.e., the current source  121 . Therefore, when the current output capacity of the transistor Q 12  is well over the current output capacity of the transistor Q 11 , most of the current provided to the current source  121  is from the transistor Q 12 . 
   As a current mirror circuit is formed by the transistors M 11  and M 12 , a current same as the collector current of the transistor Q 12  flows through the transistor Q 13 . Accordingly, a signal same as the signal appearing at the base of the transistor Q 12  appears at the base of the transistor Q 13  as shown in (B). 
   With this configuration, the maximum value output circuit  91  can output higher one of the input signals V 12  and V 13 . In addition, since bipolar transistors are used as the input/output transistors Q 11 , Q 12  and Q 13  in this embodiment, the difference between the current supply capacities of the transistors Q 11  and Q 12  based on the detection signals V 12  and V 13  is increased. Therefore, the maximum value output circuit  91  can accurately output a maximum value. 
   It should be understood, although a circuit for detecting a state of that CCFLs is exemplified in the above embodiment, the drive state detection circuit of the present invention is not limited thereto. The drive state detection circuit of the present invention is a circuit that detects a state of parts driven by alternating currents, and is applicable to other devices without being limited to the above specific examples. 
   The present application is based on Japanese Priority Application No. 2004-242470 filed on Aug. 23, 2004, with the Japanese Patent Office, the entire contents of which are hereby incorporated by reference.