Abstract:
An apparatus and method for channel count reduction in solid-state-based positron emission tomography that multiplexes read-outs from photo-detectors using a sum delay circuit, including a sum channel and a delay-sum channel. The sum channel sums signals from sensors in an array and is digitized to extract the timing and energy information. A delay-sum channel includes a discrete delay line that introduces a known delay after each sensor, creating a time signature for the sensor, followed by a summing circuit that adds the delayed signals. The delay-sum channel is digitized using a high speed counters to extract location information. Start and Stop signals for the counter are derived when the sum channel output and the delay-sum channel output cross a pulse ID threshold, respectively. The pulse ID threshold is chosen to minimize the Compton scatter and not clip the photo-peak events.

Description:
FIELD 
       [0001]    The present disclosure generally relates to an apparatus and method for channel count reduction in positron emission tomography (PET). More specifically, the present disclosure relates to an apparatus and method for channel count reduction in solid-state-based positron emission tomography, by multiplexing read-outs from photo-detectors. 
       BACKGROUND 
       [0002]    PET imaging starts with the administration (mostly injection but also inhalation and ingestion) of a radiopharmaceutical to a patient, and in time, the physical and bio-molecular properties of the agent will concentrate at specific locations in the human body. The actual spatial distribution, the intensity of the point or region of accumulation, and the kinetics of the process from administration to capture to eventually elimination, are all elements that may have a clinical significance. During this process, the positron emitter attached to the pharmaceutical agent will emit positrons according to the physical properties of the isotope (half-life, branching ratio). Each positron will eventually interact with an electron of the object to get annihilated, and produce, most of the time, two gamma rays at 511 keV at substantially 180 degrees from each other. Those two gamma rays can be detected preferably using a scintillating crystal followed by a photo-detector and processing electronics. 
         [0003]    Drawing a line between the detected locations of a pair of gamma rays, also called as a line-of-response (LoR), one can retrieve the likely location of the original annihilation event. While this process will only identify a line along which the annihilation event has occurred, by accumulating a large number of these lines, and through the tomographic reconstruction process, the original distribution of positron emitters can be estimated. 
         [0004]    In addition to the location of the two scintillation events, if accurate timing (within a few hundred picoseconds) is available, the time it takes for the gamma ray to traverse from its point of origin to the detector electronics can be precisely calculated. This time, called time-of-flight (ToF), adds more information on the likely position of its origin, viz. the annihilation location, along the line. Limitations in the timing resolution of the scanner will determine the accuracy of the positioning along this line. Limitations in the determination of the location of the original scintillation events will determine the ultimate spatial resolution of the scanner, while the specific characteristics of the isotope (energy of the positron) will also contribute, via positron range and co-linearity of the two gamma rays, to the determination of the spatial resolution for a specific agent. 
         [0005]    The above calculations need to be repeated for a large number of events. While every case needs to be analyzed to determine how many counts (paired events) are required to support the imaging tasks, current practice dictates that a typical 100 cm long, FDG (fluoro-deoxyglucose) study should accumulate hundreds of millions of counts. The time it will take to accumulate this number of counts is determined by the injected dose and the sensitivity and counting capacity of the scanner. 
         [0006]    The counting capacity of the scanner is determined mainly by two factors. First, the decay time of the scintillating crystal determines the time for which the sensor will be occupied. This is an irreducible parameter and is inherent to the very nature of the crystal. The time it takes for the scintillation light to be created and integrated will determine the minimum amount of time for event processing. If another event were to occur during this period of time, the light from the second event would bias the estimation of the event, to the point of rendering the estimation of timing, energy, and position invalid, and both events might even need to be discarded. 
         [0007]    The second variable influencing the count rate is the fraction of the total surface of the scanner “affected” or “occupied” during an event, also called a trigger zone. A second event can occur at any time during the processing of a first event if it occurs outside the trigger zone in a different location, and uses a different set of electronics channels. Consequently, the notion of trigger zones and integration time is critical in determining the counting rate capacity of a scanner. 
         [0008]    Therefore, a desire exists to create a very small trigger zone to maximize count rate. However, this is immediately opposed by the fact that a smaller detector area will also increase the electronics channel count and, in turn, the overall system cost. 
         [0009]    Conventional approaches typically implement multiplexing of channels when the channel count becomes too high to implement the signal processing electronics using discrete components. For example, a typical modern PET scanner will utilize only a few hundred sensors for the entire scanner, and one-to-one coupling of sensors to the processing electronics channel is clearly affordable in this case. 
         [0010]    The use of other types of sensors, for example, multi-anode photo-multiplier tubes (PMT), photodiodes, avalanche photodiodes (APD), or more recently, solid-state photo-multipliers (SiPM, SSPM, MPPC etc) will usually result in several thousands or even tens of thousands of sensors. In this case, efforts to optimize the channel count are clearly required. Two such channel count reduction techniques are to use summing circuits by quadrant (Anger logic), or by rows and columns, as shown in  FIGS. 1 and 2 , respectively. 
         [0011]      FIG. 1  shows a variation of Anger logic by quadrant, wherein a block of Bismuth Germanium Oxide (BGO) crystals is read out by 4 quadrants  101  created by summing 2×2 matrices of SiPM  103 . Anger logic is applied to the outputs of these quadrants to estimate the position, energy, and timing. 
         [0012]      FIG. 2  shows a conventional row-column read-out using row-wise and column-wise summing (Σ) channels. In this example case of a 4×4 matrix of 16 sensors  201  (or sensor cells), 4 rows and 4 columns can properly identify the location of the event in the array. This approach uses only 8 electronics channels compared to dedicated electronics channel for each sensor, where the channel count is 16. The larger the array, the more channel count reduction is achieved by this approach, e.g., in the case of square arrays where N is the number of sensors in every row and column, the difference between N×N and 2N becomes significant. 
         [0013]    One problem with this approach is that the signal is divided into multiple summing circuits in the case of gamma interactions from multiple crystals (via Compton scattering in the first crystal). This situation is quite common. In a typical PET scanner construction with 20 mm thick LYSO and 4×4 mm 2  crystals, only 78% of all interactions affect a single crystal, and the remaining 22% of the events produce light in at least two neighboring crystals. This proportion varies with the actual design of the crystal sizes and thicknesses, but remains a significant problem that needs to be addressed. 
         [0014]    Most of these multiple-interaction events will affect the very first row of crystals around the crystal of interest, creating a possible matrix of nine crystals including the crystal of interest, centered at the crystal of interest. If partial energy deposition occurs at the left or right of the central crystal, the horizontal line should capture the sum of the two sensors affected. If it is at the top or bottom of the central crystal, the vertical line should do the same. However, any oblique interaction will distribute the signal over multiple summing lines. 
         [0015]    The overall effect of this approach is that the system will need to analyze both horizontal and vertical lines on summed signals to find out which one contains the best signal for timing pick-off Moreover, in the case of oblique interactions, the best signal will be degraded, thereby limiting the ability of a good timing estimation. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0016]    A more complete appreciation of the embodiments described herein, and many of the attendant advantages thereof will be readily obtained as the same becomes better understood by reference to the following detailed description when considered in connection with the accompanying drawings, wherein: 
           [0017]      FIG. 1  shows a variation of Anger logic by quadrant, where a block of BGO is read out by 4 quadrants created by summing 2×2 SiPM matrices; 
           [0018]      FIG. 2  shows a conventional row-column read-out using row-wise and column-wise summing (Σ) channels; 
           [0019]      FIG. 3  shows a schematic diagram of Σ and ΣΔ channel multiplexing for an array of 4 sensors; 
           [0020]      FIGS. 4A and 4B  show the sensor outputs for an array of 4 sensors before and after being selectively delayed, respectively; 
           [0021]      FIG. 5  shows a block diagram of Σ and ΣΔ channels; 
           [0022]      FIG. 6  shows the timing diagrams of the Σ and ΣΔ channels; 
           [0023]      FIG. 7  shows the intensity function for a BGO scintillator used to determine photon arrival times; 
           [0024]      FIGS. 8A and 8B  show arrival times, which represent illustration of typical paths taken by a non-homogenous Poisson process, and typical current pulse out of the BGO-PMT setup using a superposition model, respectively; 
           [0025]      FIG. 9  is a flowchart of a simulation setup used to study the ΣΔ scheme; 
           [0026]      FIGS. 10A-10D  show the results of a ΣΔ Monte Carlo simulation; 
           [0027]      FIG. 11  shows a flowchart for a method of data acquisition in a gamma ray detector; and 
           [0028]      FIG. 12  shows a block diagram of a gamma ray detector. 
       
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       [0029]    The present disclosure describes a method to reduce the channel count while allowing the use of a smaller trigger zone and affording a very high count rate capacity and greater positioning accuracy. Disclosed embodiments offer a greater channel count reduction, while also providing a more optimal system design, with one part of the circuitry dedicated to timing and energy estimation, and another part to positioning into the array. 
         [0030]    According to a first embodiment, there is provided a data acquisition device for a gamma ray detector, comprising a summing circuit configured to sum a plurality of electrical signals from a corresponding plurality of sensors coupled to an array of scintillation crystals to generate a first signal, the plurality of sensors converting received light into the plurality of electrical signals, wherein the light is generated by a crystal of interaction in response to incident gamma rays generated by an annihilation event, a delay summing circuit configured to selectively delay and sum the plurality of electrical signals to generate a second signal, a first circuit configured to receive the first signal and to determine an energy and an event time of the first signal, and a second circuit configured to receive the first signal and the second signal and to determine, when the first signal exceeds a predetermined threshold, which sensor of the plurality of sensors corresponds to a location of the crystal of interaction. 
         [0031]    According to a second embodiment, there is provided a gamma ray detector, comprising an array of scintillation crystals, an array of sensors, and a plurality of data acquisition devices, each data acquisition device including a summing circuit configured to sum a plurality of electrical signals from a corresponding plurality of sensors coupled to the array of scintillation crystals to generate a first signal, the plurality of sensors converting received light into the plurality of electrical signals, wherein the light is generated by a crystal of interaction in response to incident gamma rays generated by an annihilation event, a delay summing circuit configured to selectively delay and sum the plurality of electrical signals to generate a second signal, a first circuit configured to receive the first signal and to determine an energy and an event time of the first signal, and a second circuit configured to receive the first signal and the second signal and to determine, when the first signal exceeds a predetermined threshold, which sensor of the plurality of sensors corresponds to a location of the crystal of interaction. 
         [0032]    According to a third embodiment, there is provided a gamma ray detection method, comprising generating a first signal by summing a plurality of electrical signals from a corresponding plurality of sensors coupled to an array of scintillation crystals, the plurality of sensors converting received light into the plurality of electrical signals, wherein the light is generated by a crystal of interaction in response to incident gamma rays generated by an annihilation event, generating a second signal by selectively delaying and summing the plurality of electrical signals, determining an energy and an event time of the first signal, and determining, based on the second signal, which sensor of the plurality of sensors corresponds to a location of the crystal of interaction, when the first signal exceeds a predetermined threshold. 
         [0033]    Referring now to the drawings, wherein like reference numerals designate identical or corresponding parts throughout the several views,  FIG. 3  shows one embodiment using an array of sensors  301 . The four sensors shown represent a row, column, or area of the array. In this example, four sensors are shown, but any number of sensors can be used. A concept of the ΣΔ approach is to create two channels in the following fashion. The first channel is called the Σ channel, which is the output of a summation circuit  303  that sums signals from all the sensors  301  along a row (or a column), and therefore creates an optimal signal for energy estimation and timing pickoff. The second channel is called the ΣΔ channel, which is created by inserting a known delay element Δ  305  for each sensor  301 , thus creating a time signature for the location of the event, and then summing the delayed signals using a summation circuit  303 . 
         [0034]      FIGS. 4A and 4B  show an example of four sensor signals before and after being delayed, respectively. In this example, the different delay times introduced to different sensor signals are integer multiples of a predetermined delay time Δ. 
         [0035]      FIG. 5  shows a block diagram of the Σ and ΣΔ channels. Each of the Σ and ΣΔ channels are conditioned by a corresponding pre-amplifier and shaper  501 , the output of which is connected to the positive input of a corresponding comparator  505 . The negative input of the comparators  505  is connected to the output of a pulse ID threshold DAC  503 . The outputs of the comparators  505  corresponding to the Σ and ΣΔ channels provide a “start” and a “stop” signal, respectively, to a high speed counter  509 . 
         [0036]    The output signal of the pre-amplifier and shaper  501  corresponding to the Σ channel is further connected to an ADC circuit  507 , which digitizes the signal and extracts timing and energy. 
         [0037]    The ΣΔ channel is digitized by the high speed counter  509  to extract location information. (By comparison, for row-column logic, 16 channels will be digitized directly using analog-to-digital converters (ADCs), and the comparators and DACs are not required. This requires twice the number of ADCs, and in turn twice the power dissipation). 
         [0038]    The START signal is generated when the Σ channel output crosses a pulse ID threshold. In the ΣΔ channel, the signals from the SiPMs will propagate through a number of discrete delay elements with a precise time delay of Δ, and a summing Σ circuit. The pulse identification threshold in this channel provides a STOP signal. The time difference between the derived START and STOP signals provides location information on the detector line. The total delay introduced due to the delay line is less than 80 ns total for a delay chain containing 8 SiPMs. Compared to the row-column and quad multiplexing approaches, the electronic dead time per channel is greater, but due to smaller detector occupancy, its overall effect is minimal. 
         [0039]      FIG. 6  shows the timing diagrams of the Σ channel, the ΣΔ channel, and the comparator outputs. In this example, the Σ channel and the ΣΔ channel provide the sum and the delay-sum of the output signals from 4 detectors, respectively. In this example, the output signals from  4  detectors are delayed by different delay amounts, i.e., Δ, 2Δ, 3Δ, and 4Δ. A pre-calibrated Pulse ID threshold is applied to both the Σ channel and the ΣΔ channel, to extract a START signal and a STOP signal, respectively. The Pulse ID threshold is chosen such that the STOP signal corresponds to the delay time of the strongest signal among the  4  detectors. In this example, the strongest signal is delayed by 3Δ, and thus the time difference between the START and STOP signals is approximately 3Δ. 
         [0040]    The pulse ID (PID) threshold for the ΣΔ channel needs to be chosen carefully to minimize the Compton scatter and not clip the photo-peak events. Based on the energy resolution observed during measurements, the Compton valley lies around 3σ ER  lower than the photo-peak, where 2.34*σ ER  is the energy resolution (18%˜25% measured). Since energy measurement is crude and does not use integrated values from the complete scintillator pulse, placing the PID threshold at this value is prone to slightly larger Compton acceptance. In addition the noise present at the PID comparator puts a constraint on the threshold. A careful comparator design is used to minimize Compton acceptance. In order that the comparator won&#39;t trigger due to photon-statistical noise, the comparator is designed using negative hysteresis that is equal to at least 2σ PID , where σ PID  is the noise at the PID comparator input. The energy measurement error σ PID  is given by equation (1): 
         [0000]    
       
         
           
             
               
                 
                   
                     σ 
                     PID 
                   
                   = 
                   
                     
                       
                         
                           σ 
                           photon 
                           2 
                         
                          
                         
                           ( 
                           
                             t 
                             PID 
                           
                           ) 
                         
                       
                       + 
                       
                         σ 
                         ENC 
                         2 
                       
                     
                   
                 
               
               
                 
                   ( 
                   1 
                   ) 
                 
               
             
             
               
                 
                   
                     σ 
                     photon 
                     2 
                   
                   = 
                   
                     
                       qFM 
                       2 
                     
                      
                     
                       
                         ∫ 
                         
                           - 
                           ∞ 
                         
                         ∞ 
                       
                        
                       
                         
                           
                             I 
                             photon 
                           
                            
                           
                             ( 
                             α 
                             ) 
                           
                         
                          
                         
                           
                             ϖ 
                             2 
                           
                            
                           
                             ( 
                             
                               t 
                               - 
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         [0000]    where σ photon  is the photon-statistical noise, t PID  is the time when the PID threshold is crossed and σ ENC  is the electronic noise. In equation (2), q is the charge, F is an excess noise factor, M is the gain of the photo-detector, and ω is the weighting function of the preamplifier and shaper. I photon  is the intensity function for BGO and can be approximated to a double (or triple with 60 ns˜10% and 300 ns˜90%) decay exponential. 
         [0041]    The purpose of the pulse identification circuit is to locate the crystal of highest energy interaction point. Timing is extracted by applying a timing threshold to the timing comparator, which provides the START signal. To find the highest energy interaction point, the ΣΔ pulse is applied to one input of a comparator, and the second input of the comparator is set using a DAC to a pulse ID threshold value. The comparator fires giving a STOP signal when the pulse identification threshold is crossed by the ΣΔ pulse. An optimum pulse ID threshold needs to be placed at (3σ ER +3σ PID ) below the photo-peak (where σ ER  is the standard deviation of the photo-peak distribution) to minimize or reject Compton scatter and accept only the photo-peak events. 
         [0042]    Assuming a yield of 10600 photons/MeV for BGO, and 25% energy resolution, the PID threshold will be placed at or about 3680 photons. Assuming that we use a peaking time of 900 ns (a tradeoff between dead time and energy resolution will drive this number), the PID threshold will be crossed at about 600 ns. The photon noise contribution is approximately ˜60 photons at the PID comparator threshold. This would correspond to 1% of photo-peak events. Optimum SNR is obtained using an integration capacitor equal to that of the detector, the SPM35CN device capacitance (Cdet). Using these values to calculate the slope of the signal at the PID threshold crossing, and using a PDE of 10% and a gain of 3.2×10 6 , the photon statistics noise is σphoton=50 mV RMS for σ ENC =3.5 mV. 
         [0043]    The value of Δ is at least 3σ ENC  greater than the electronic noise σ ENC . Larger values of Δ are not advised because this would increase the photon-statistical noise contribution, which would otherwise cancel out, given that ΣΔ is a delayed sum version of the signal in the Σ channel. If the photon-statistical noise contribution to the START signal from a given detector element n at PID threshold crossing of Σ is σ PID n, then the photon-statistical noise contribution to the STOP signal from the delayed signal from n is simply σ PID n+δn. The overall noise contribution to START-STOP is then δn, which varies with Δ. Thus, δn will increase with a larger value of Δ, and the larger the Δ, the larger the noise contribution, since as Δ→0, photon statistic noise in (STOP-START)→0. The position information is given by the difference between the STOP and START, and the error in the position information is only limited by the non-coherent electronic noise, and not the photon statistics. Also, smaller values of Δ are advised with a theoretical lower limit given by electronic noise, σ ENC . 
         [0044]    With an appropriate choice of Δ, careful comparator design, and calibration of PID thresholds, this approach is limited only by the energy resolution, the accuracy of the pulse identification circuit, and the high-speed counter frequency. The overall number of delays that the system can include, and thereby the size of the array that can be multiplexed, is roughly dependent on the processing time for one event, as the energy, timing, and position information will eventually need to come together to define an event. Encoding of fast crystals such as LYSO, with ˜40 ns decay time may be limited to 4×10 ns individual delays, assuming 10 ns as a reasonable delay quantum, and the same delay chain on a slower crystal like BGO, with 300 ns decay, would allow more than 30 sensors to be encoded. For an 8×8 matrix, this would entail an additional 80 ns of dead time on top of the chosen integration time, compared to the row-column logic concept, which is a small price to pay for a smaller trigger zone area and the channel count reduction that this embodiment offers. 
         [0045]    The BGO timing resolution is poorer than the LYSO. The first photo-electron timing errors for two different energy discrimination thresholds are shown in  FIG. 6 . The source of this timing error is the non-homogeneous Poissonian nature of the photon arrivals at the face of the photo-detector. For BGO, due to lower number of generated photons stretched on a larger time scale of 300 ns (90%), such a fluctuation is observed. 
         [0046]    The RMS error in the timing measurement is dependent on the threshold as well as the photon collection. From the literature, the error lies between 2 ns to 7 ns for a given threshold. This would dictate the coincidence timing window 2τ using the BGO scintillator, and would be about 2.8 ns˜9.8 ns. 
         [0047]      FIG. 7  shows the intensity function for a BGO scintillator used to determine photon arrival times. Matlab simulation of a non-homegenous Poissionian process with a double-decay exponential model of BGO scintillator response was used to obtain this function. 
         [0048]      FIGS. 8A and 8B  show the arrival times that represent an illustration of a typical path taken by a non-homogenous Poisson process, and a typical current pulse out of the BGO-SiPM setup using a superposition model, respectively. The BGO decay model is convolved with the SiPM single electron response and a randomly picked gain value from the independent identically distributed PDF. The pulses seen near the floor are an illustration of a single photon response. 
         [0049]    A similar simulation study for the BGO scanner with an average SiPM response was conducted. The flowchart in  FIG. 9  summarizes the setup used for the study. Starting with a mean 10600 photons/MeV, the signal chain is simulated to reflect the photon detection efficiency, energy blurring, the BGO response intensity function shown in  FIG. 8 , and non-homogenous Poissonian arrival times of the photons. These are then convolved with the single SiPM cell response, which approximates the SiPM output. Monte Carlo simulation is performed to simulate 100 gammas. Randomly selected photo-peak and lower energy events are used as input to the ΣΔ operator. The outputs of the Σ and ΣΔ channels are then convolved with the second order preamp shaper impulse response. As described earlier, pulse ID threshold is placed such that Compton scatter is rejected. The pulse is identified when the preamp-shaper output crosses the PID threshold. The START time is obtained when the Σ channel crosses the PID threshold. Similarly, the time instance when ΣΔ channel crosses the same PID threshold is labeled as the STOP time. The PID is then given by equation (3). 
         [0000]    
       
         
           
             
               
                 
                   PID 
                   = 
                   
                      
                     
                       
                         Stop 
                          
                         
                           - 
                         
                          
                         Start 
                       
                       Δ 
                     
                      
                   
                 
               
               
                 
                   ( 
                   3 
                   ) 
                 
               
             
           
         
       
     
         [0050]    Monte Carlo simulations for 25 different combinations of photo-peak and scatter were performed with 5 different threshold values.  FIGS. 10A-10D  show the results of these ΣΔ Monte Carlo simulations. In each case, the photo-peak pulse was in location 3 with two Compton neighbors. The PID formula was able to identify the pulse every single time without an error. The standard deviation in the START and STOP signals was in the range of 19 ns to 22 ns, as expected. Also, as anticipated from the theory described above, the standard deviation in the START-STOP ranged from 180 ps to 260 ps. The value of Δ=10 ns was able to successfully decode the PID location, as seen in  FIGS. 10A-10D . 
         [0051]    Embodiments disclosed herein offer a factor of 4 reduction in the number of electronics channels. The embodiments also afford a very small trigger zone compared to conventional approaches. The count rate performance for a scanner with such a readout scheme would be superior to the conventional approaches, and channel count reduction will be equivalent to the Anger logic method. Inter-crystal scatter causes mispositioning of scintillation events. The ability and limits of ΣΔ to handle multi-crystal events are better than the conventional approaches. Just by its design, ΣΔ adapts the highest energy positioning method to locate multi-crystal events. Compared to the Anger logic method, which uses a weighted-energy positioning method, the ΣΔ method offers 10% or better positioning accuracy. This is done by plainly rejecting low-energy scatters by employing the calibration method to identify the scatter threshold. 
         [0052]    Embodiments disclosed herein utilize discrete commercial off-the-shelf components. Alternative embodiments of this scheme will implement an Application Specific Integrated Circuit (ASIC) in a standard complementary metal-oxide semiconductor (CMOS) technology, affording much smaller real estate and low power consumption. 
         [0053]    Embodiments disclosed herein will provide a maximum channel count reduction at a very high count rate performance, therefore a better cost structure, while maintaining a high fidelity of the sensor responses. In some cases, it is believed that a 2-channel system by daisy chaining the rows in order to create Σ and ΣΔ channels may harness a 16-cell sensor matrix, while the conventional row-column system would require 8 channels. 
         [0054]      FIG. 11  shows a flowchart for a method of data acquisition in a gamma ray detector. The flowchart includes one process including steps S 1101 , S 1105 , and S 1107 , and another process including steps S 1103 , S 1109 , S 1111 , S 1113 , and S 1115 . 
         [0055]    In step S 1101 , a sum signal of the electrical signals from sensors coupled to an array of scintillation crystals is generated. 
         [0056]    In step S 1103 , a delay sum signal of these electrical signals is generated such that each signal is delayed by a different amount of time. 
         [0057]    In step S 1105  the sum signal is compared with a pulse ID threshold, and if the sum signal is larger than the pulse ID threshold, a counter is started in step S 1107 . Otherwise, the process goes back to step S 1101 . 
         [0058]    In step S 1109  the delay sum signal is compared with the pulse ID threshold, and if the delay sum signal is larger than the pulse ID threshold, the counter is stopped in step S 1111 . Otherwise, the process goes back to step S 1103 . 
         [0059]    In step S 1113  the location of an event is calculated. 
         [0060]    In step S 1115  the timing and energy of the event is determined. 
         [0061]      FIG. 12  shows an exemplary hardware configuration that can be used with the present technological advancement to detect gamma rays. In  FIG. 12 , photomultiplier tubes  135  and  140  are arranged over light guide  130 , and the array of scintillation crystals  105  is arranged beneath the light guide  130 . A second array of scintillation crystals  125  is disposed opposite the scintillation crystals  105  with light guide  115  and photomultiplier tubes  195  and  110  arranged thereover. 
         [0062]    In  FIG. 12 , when gamma rays are emitted from a body under test (not shown), the gamma rays travel in opposite directions, approximately 180° from each other. Gamma ray detection occurs simultaneously at scintillation crystals  100  and  120 , and a scintillation event is determined when the gamma rays are detected at scintillation crystals  100  and  120  within a predefined time limit. Thus, the gamma ray timing detection system detects gamma rays simultaneously at scintillation crystals  100  and  120 . However, for simplicity only, gamma ray detection is described relative to scintillation crystal  100 . One of ordinary skill in the art will recognize, however, that the description given herein with respect to scintillation crystal  100  is equally applicable to gamma ray detection at scintillation crystal  120 . 
         [0063]    Each photomultiplier tube  110 ,  135 ,  140  and  195  is respectively connected to data acquisition unit  150 . The data acquisition unit  150  includes hardware configured to process the signals from the photomultiplier tubes. The data acquisition unit  150  measures the arrival time of the gamma ray. The data acquisition unit  150  produces two outputs (one for the combination of PMT  135 / 140  and one for the combination of PMT  110 / 195 ) which encodes the time of the discriminator pulse relative to a system clock (not shown). For a time of flight PET system, the data acquisition unit  150  typically produces a time stamp with an accuracy of 15 to 25 ps. The data acquisition unit measures the amplitude of the signal on each PMT (four of the outputs from data acquisition unit  150 ). 
         [0064]    The data acquisition unit outputs are provided to a CPU,  170 , for processing. The processing consists of estimating an energy and position from the data acquisition unit outputs and an arrival time from the time stamps output for each event, and may include the application of many correction steps, based on prior calibrations, to improve the accuracy of the energy, position, and time estimates. As one of ordinary skill in the art would recognize, the CPU  170  can be implemented as discrete logic gates, as an Application Specific Integrated Circuit (ASIC), a Field Programmable Gate Array (FPGA) or other Complex Programmable Logic Device (CPLD). An FPGA or CPLD implementation may be coded in VHDL, Verilog or any other hardware description language and the code may be stored in an electronic memory directly within the FPGA or CPLD, or as a separate electronic memory. 
         [0065]    Further, the electronic memory may be non-volatile, such as ROM, EPROM, EEPROM or FLASH memory. The electronic memory may also be volatile, such as static or dynamic RAM, and a processor, such as a microcontroller or microprocessor, may be provided to manage the electronic memory as well as the interaction between the FPGA or CPLD and the electronic memory. 
         [0066]    Alternatively, the CPU  170  may be implemented as a set of computer-readable instructions stored in any of the above-described electronic memories and/or a hard disc drive, CD, DVD, FLASH drive or any other known storage media. Further, the computer-readable instructions may be provided as a utility application, background daemon, or component of an operating system, or combination thereof, executing in conjunction with a processor, such as a Xenon processor from Intel of America or an Opteron processor from AMD of America and an operating system, such as Microsoft VISTA, UNIX, Solaris, LINUX, Apple, MAC-OS and other operating systems known to those skilled in the art. 
         [0067]    Once processed by the CPU  170 , the processed signals are stored in electronic storage  180 , and/or displayed on display  145 . As one of ordinary skill in the art would recognize, electronic storage  180  may be a hard disk drive, CD-ROM drive, DVD drive, FLASH drive, RAM, ROM or any other electronic storage known in the art. Display  145  may be implemented as an LCD display, CRT display, plasma display, OLED, LED or any other display known in the art. As such, the descriptions of the electronic storage  180  and the display  145  provided herein are merely exemplary and in no way limit the scope of the present advancements. 
         [0068]      FIG. 12  also includes an interface  175  through which the gamma ray detection system interfaces with other external devices and/or a user. For example, interface  175  may be a USB interface, PCMCIA interface, Ethernet interface or any other interface known in the art. Interface  175  may also be wired or wireless and may include a keyboard and/or mouse or other human interface devices known in the art for interacting with a user. 
         [0069]    In the above description, any processes, descriptions or blocks in flowcharts should be understood as representing modules, segments or portions of code which include one or more executable instructions for implementing specific logical functions or steps in the process, and alternate implementations are included within the scope of the exemplary embodiments of the present advancements in which functions may be executed out of order from that shown or discussed, including substantially concurrently or in reverse order, depending upon the functionality involved, as would be understood by those skilled in the art. 
         [0070]    While certain embodiments have been described, these embodiments have been presented by way of example only, and are not intended to limit the scope of the inventions. Indeed, the novel methods, apparatuses and systems described herein may be embodied in a variety of other forms; furthermore, various omissions, substitutions and changes in the form of the methods, apparatuses and systems described herein may be made without departing from the spirit of the inventions. The accompanying claims and their equivalents are intended to cover such forms or modifications as would fall within the scope and spirit of the inventions.