Abstract:
Described are regulated cascode amplifiers with improved low-voltage performance. The improved amplifier is similar to conventional regulated cascode amplifiers, including a cascode circuit and a feedback amplifier. The cascode circuit conventionally includes two output transistors, the first of which preferably remains in saturation to provide a relatively stable output resistance over a range of output voltages. A booster circuit in accordance with one embodiment maintains the first transistor of the cascode circuit in saturation over a broader range of output voltages, and consequently extends the low-end of the operating range of the cascode amplifier.

Description:
BACKGROUND 
     FIG. 1  (prior art) depicts a regulated cascode (RGC) amplifier  100 , which has a relatively high output impedance r 0  and a wide output voltage swing. RGC  100  includes a feedback amplifier  105 , a bias voltage terminal V BIAS , a cascode circuit  110 , an input terminal V IN , and an output terminal V OUT . Feedback amplifier a  105  includes a first current sourcing transistor  120  and amplifier transistor  125  connected in series between first and second power supply terminals V DD  and V SS . Cascode circuit  110  includes input and output transistors  130  and  135  connected in series between output terminal V OUT  and supply terminal V SS . 
   In operation, input transistor  130  converts input voltage V IN  into a drain current I 0  that flows through the drain-source path of output transistor  135  to output terminal V OUT . The drain-source voltage across transistor  130  should be relatively stable to suppress channel-length modulation that might otherwise reduce output impedance r 0 . The drain-source voltage of transistor  130  is therefore regulated about a fixed value by a feedback loop that includes amplifier transistor  125  and output transistor  135 . Feedback amplifier  105  stabilizes the drain-source voltage of transistor  135  even when transistor  135  is biased in the linear region, which extends the usable range of the output signal V OUT . 
     FIG. 2  (prior art) depicts an improved regulated cascode amplifier (IRGC)  200 . IRGC  200  is similar to a regulated cascode circuit  100  of  FIG. 1 , similar components having the same label and function. In addition to feedback amplifier circuit  105  and cascode circuit  110 , IRGC  200  includes a level shifter  205 . Level shifter  205  in turn includes a diode-connected transistor  210  and a second current-sourcing transistor  215  connected in series between supply terminal V DD  and the drain of transistor  130 . The gate and drain of transistor  215  connect to bias voltage terminal V BIAS  and the drain of transistor  210 , respectively. The gate and drain of transistor  210  connect to the gate of transistor  125 , while the source connects to the drain of transistor  130 . 
   The inclusion of level shifter  205  provides improved performance for low-voltage applications. Level shifter  205  limits the drain-source voltage V DS130  of transistor  130  to the difference between the gate-source voltage V GS125  of transistor  125  and the gate-source voltage V GS210  of transistor  210  (i.e., V DS130 =V GS125 −V GS210 ). This relatively low voltage at the drain of transistor  130  reduces the minimum level for output voltage V OUT . 
   The performance of IRGC  200  depends to a large extent on the characteristics of feedback amplifier  105 , which in turn depends on the transconductance g M  of transistor  125 . A high g M , obtained by increasing the width W of transistor  125 , improves the response time of feedback amplifier  105 , a desirable characteristic for high-speed circuits. Increasing the width also reduces the gate-source voltage V GS125  of transistor  125 , and consequently the drain-source voltage V DS130  across transistor  130 . The relationship between the width of transistor  125  and the drain-source voltage V DS130  of transistor  130  sets an upper limit on the width of transistor  125 : if the width of transistor  125  is too high, the drain-source voltage V DS130  of transistor  130  can be reduced to levels that bring transistor  130  into the linear range. This is undesirable, as the output resistance r 0  of IRGC  200  varies considerably with output voltage V OUT  when transistor  130  operates in the linear region. Unfortunately, the constraints on the width of transistor  125  limit the speed performance of IRCC  200  in low-voltage applications. 
   SUMMARY 
   The present invention is directed to a regulated cascode amplifier with improved low-voltage performance. The improved amplifier is similar to conventional regulated cascode amplifiers, including a cascode circuit and a feedback amplifier. The cascode circuit conventionally includes two output transistors, the first of which preferably remains in saturation to provide a relatively stable output resistance over a range of output voltages. A booster circuit in accordance with one embodiment maintains the first transistor of the cascode circuit in saturation over a broader range of output voltages, and consequently extends the low-end of the operating range of the cascode amplifier. 
   This summary does not limit the invention, which is instead defined by the claims. 

   
     BRIEF DESCRIPTION OF THE FIGURES 
       FIG. 1  (prior art) depicts a simple regulated cascode amplifier. 
       FIG. 2  (prior art) depicts an improved regulated cascode amplifier having a level-shifter. 
       FIG. 3A  depicts an embodiment of a boosted regulated cascode amplifier. 
       FIG. 3B  depicts another embodiment of a boosted regulated cascode amplifier. 
       FIG. 4  depicts an embodiment of a current mirror  400  that employs boosted regulated cascode amplifier of  FIG. 3A . 
       FIG. 5  depicts a current mirror  500  in accordance with another embodiment. 
   

   DETAILED DESCRIPTION 
     FIG. 3A  depicts an amplifier  300  in accordance with one embodiment of the present invention. Amplifier  300  is similar to improved regulated cascode amplifier  200  of  FIG. 2 , similar components having the same label and function. Amplifier  300  is modified, however, to reduce the constraints on transistor  125  and consequently to increase output impedance r 0  and speed performance in low-voltage applications. Though amplifiers  300  and  200  are similar, various design parameters, such as transistor aspect ratios (W/L), will vary as necessary to adapt amplifiers  300  and  200  to particular applications, as will be understood by those of skill in the art. 
   In common with amplifier  200  of  FIG. 2 , amplifier  300  includes feedback amplifier  105 , cascode circuit  110 , and level shifter  205 . Amplifier  300  additionally includes a booster transistor  305  connected between feedback amplifier  105  and ground potential GND. The drain and gate of booster transistor  305  connect to the source (feedback-amplifier terminal) and gate (feedback-amplifier bias terminal) of amplifier transistor  125 , respectively. 
   Applying Kirchoff&#39;s Voltage Law (KVL) to a voltage loop formed by transistors  305 ,  125 ,  210 , and  130  elucidates the significance of booster transistor  305 . Starting at the source of input transistor  130  and traversing the voltage loop counterclockwise, the loop KVL equation is:
 
 V   DS130   +V   TH210   +ΔV   210   −V   TH125 −ΔV 125   −V   DS305 =0  (1)
 
where V DS130  is the drain-source voltage of input transistor  130 ; V DS305  is the drain-source voltage of booster transistor  305  and is equal to a booster voltage V BT ; V TH125  is the threshold voltage of amplifier transistor  125 ; ΔV 125  is an excess voltage across amplifier transistor  125  necessary for transistor  125  to sink current sourced by transistor  120 ; V TH210  is the threshold voltage of diode-connected transistor  210 , and ΔV 210  is an excess voltage across transistor  210  necessary for transistor  210  to sink current sourced by transistor  215 .
 
   Substituting booster voltage V BT  for drain-source voltage V DS305  of transistor  305  and making drain-source voltage V DS130  of input transistor  130  the subject of equation 1, equation 1 then can be simplified to:
 
 V   DS130   =V   BT   +ΔV   125   +V   TH125   −V   TH210   −ΔV   210   (2)
 
Assuming that the threshold voltages V TH125  and V TH210  and the excess voltages ΔV 210  and ΔV 125  of respective transistors  125  and  210  are equal simplifies Equation 2 to:
 
V DS130 =V BT   (3)
 
Equation 3 is the basis of designing amplifier  300  to reduce or eliminate the limitations of amplifier  200  of  FIG. 2 . From equation 3, it can be seen that booster circuit  305  can be used to set a minimum voltage on the drain of input transistor  130 . Setting booster voltage V BT  equal to the saturation voltage V DSsat130  of input transistor  130  prevents transistor  130  from drifting out of the saturation mode.
 
   In one embodiment, transistors  125  and  210  are matched to reduce equation 2 to equation 3. The components of booster transistor  305  can also be selected so that when amplifier  300  is biased as depicted in  FIG. 3A , booster voltage V BT  generally coincides with saturation voltage V DSsat130 . 
     FIG. 3B  depicts an amplifier  310  in accordance with another embodiment of the present invention. Amplifier  310  is similar to amplifier of  FIG. 3B  and the same elements are labeled with the same numbers. In amplifier  310  booster circuit  305  is replaced by a resistor  307  connected between the source of transistor  125  and the second supply terminal V SS . Resistor  307 , i.e., a bias circuit, provides the bias voltage (i.e., booster voltage V BT ) for current source transistor  130  to maintain transistor  130  in saturation mode. The bias current for transistor  130  is provided by the current mirror transistor  215  to transistor  120 . The effect of process variation on the resistor  307  may be compensated for by making the bias current a function of the type of resistor. At least one advantage of having resistor  307  is that the size restrictions on transistors  125  and  210  are relaxed compared to the amplifier circuit without resistor  307 . 
     FIG. 4  depicts a current mirror  400  that includes an embodiment of amplifier  300  of  FIG. 3A  (or of amplifier  310  of  FIG. 3B ) and a bias circuit  405 . Multiplier  400  additionally includes a reference terminal connected to a bias terminal V BIAS  of amplifier  300 . The reference terminal receives a reference current I REF  and generates a bias voltage V BIAS . Bias circuit  405  includes a reference transistor  410  connected between power supply terminal V DD  and a bias terminal V BIAS , and a series of transistors  415 ,  420 , and  425  connected between supply terminals VDD and GND. Bias circuit  405  additionally includes a pair of transistors  430  and  435  connected between power supply terminals V DD  and GND. 
   During operation, bias voltage V BIAS  biases diode-connected transistor  410  to source a reference current I REF , which is then mirrored by mirror transistors  435 ,  425 ,  120 , and  215 . The resulting currents I 435 , I 425 , I 120 , and I 215  are proportional to reference current I REF . As is conventional in current multiplier circuits, the proportionality of a mirrored current to the reference current is determined by the ratio of the aspect ratios of the mirror and reference transistors, so the value of each of currents I 435 , I 425 , I 120 , and I 215  depends on the aspect ratio of respective transistors  435 ,  425 ,  120  and  210 . 
   Current I 435  establishes a first bias voltage V B435  on the drain and gate of diode-connected transistor  430  and the gate of transistor  420 . Diode-connected transistor  430  ensures that voltage V B435  never falls below the sum of the threshold and excess voltages of transistor  430  (i.e., V th430 +ΔV 430 ) above ground potential GND to keep transistors  420  biased on. Similarly, current I 425  establishes a second bias voltage V IN  at the drain of transistor  420 , the gate of transistor  415 , and the input terminal V IN  of amplifier  300 . To provide the best match between reference current I REF  and output current i 0 , the drain-source voltage V DS415  of transistor  415  should match the drain-source voltage V DS130  of input transistor  130 . Transistor  420  is therefore included to more closely match the drain-source voltages of transistors  415  and  130 . 
   Recalling from the discussion of regulated cascode 100 of  FIG. 1 , input transistor  130  converts input voltage V IN  into an output current i 0 . It is important to maintain input voltage V IN  at a fixed value to bias input transistor  130  to sink a constant output current i 0 . Input voltage V IN  is therefore stabilized by biasing all the transistors of bias circuit  405  in the saturation mode and employing a positive feedback mechanism to regulate input voltage VIN. Transistors  420  and  415  provide the positive feedback by sinking more current I 425  when input voltage V IN  increases and limiting current I 425  when input voltage V IN  decreases. When voltage V IN  increases, the drain-source voltage of transistors  420  and  425  likewise increase. However, because transistor  420  is already in saturation, the increase in drain-source voltage has little effect on the current through transistor  420 . On the other hand, increased gate-source voltage strongly affects transistor  415  by decreasing its drain voltage. Transistor  420  transmits the reduction in the drain voltage of transistor  415  to terminal V IN , which counteracts the increase in input voltage V IN . A reduction of input voltage V IN  triggers the opposite response. Thus by tightly regulating input voltage V IN  about a fixed value and taking advantage of the superior characteristics of amplifier  300 , current multiplier  400  generates an output current i 0  that is generally a precise multiple of reference current I REF . 
   Table 1 below shows some typical sizes of some transistors of current multiplier  400  in one embodiment. 
                                                                 TABLE 1               TRANSISTOR   410   425   415   210   130                                WIDTH (W)   11.27   112.7   23.996   311.948   240.240       LENGTH   0.805   0.805   0.28   0.28   0.126       (L)       I REF     1   100   100   1300   1300       MULTIPLIER                    
One embodiment of current multiplier  400  sized as shown in Table 1 provides an output current approximately 1300 times the reference current.
 
     FIG. 5  depicts a current mirror  500  in accordance with another embodiment. Like current mirror  400  of  FIG. 4 , current mirror  500  includes an embodiment of amplifier  300  of  FIG. 3A . In place of bias circuit  405 , however, current mirror  500  includes a bias circuit  505 . Bias circuit  505  is a mirror image of amplifier  300 , like-numbered elements being the same or similar. An external reference current Iref is driven into the node of bias circuit  505  that is analogous to the output node V OUT  of amplifier  300 . The similarity between bias circuit  505  and amplifier  300  is advantageous because both circuits will track closely in response to process, voltage, and temperature variations. In other embodiments, the transistors in amplifier  300  are sized differently than corresponding transistors in bias circuit  505  to produce a current multiplier. 
   While the present invention has been described in connection with specific embodiments, variations of these embodiments will be obvious to those of ordinary skill in the art. For example, in embodiments such as those depicted in  FIGS. 4 and 5  a single bias circuit can provide biasing voltages for a number of amplifiers  300  to produce a plurality of similar or different output currents. Therefore, the spirit and scope of the appended claims should not be limited to the foregoing description.