Abstract:
Noise gate control circuitry eliminates in audio systems the effects of spurious signals on a transducer of a speaker. The noise gate control circuitry coordinates the electronic switching of the transducer to power sources in accordance with a remote turn on control signal, and coordinates this action with the application of power to power sources and signal processing circuitry via remote turn on provisions. The noise gate control circuitry senses and responds to the signals and rapidly connects and disconnects the transducer when the signal level drops to a level too low to be of any good effect. The effect of signal level on delays is removed to provide for suppressing undesirable changes in response.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to noise gate control circuitry for electronic systems. More particularly, the present invention relates to noise gate control circuitry for audio systems which eliminates the turn on and turn off effects on speakers. 
     2. Description of the Related Art 
     It is often the case when electronic systems are able to render intended disturbances, that there are times when the rendering of unintended disturbances is affected such as when power is applied or removed from the electronic system, or at times when no disturbing effects are desired. For example, a car stereo system produces a disturbance via electronic circuitry that human hearing responds to. However, any disturbance not related to the desired sound program can be vexing to the listener. For example, sounds often occur when the system is turned on and off, and often johnston hiss noises and engine noises can be heard through the speakers when the program signal level drops low enough. 
     Currently, devices exist which interrupt or attenuate signals when their signal levels drop below given thresholds. In this way, when the program source signal drops to a level low enough that such undesirable noises dominate, the signal is automatically muted or attenuated. However, none of these devices act on the electrical connection between the transducer of a speaker and the source of electrical excitation. Typically these devices respond to the signal level of the source signal after it has been averaged by a low pass filter. Upon responding, some of these devices shift the level at which their state of response changes so as to delay further response until significant enough changes in level for a long enough period of time are sensed. These delays vary with the level of the signal and thus often compromise the desired results. 
     Devices also exist which connect transducers to a source of electrical excitation only after a delay when a remote power control signal is detected. In this way any turn on transients that might exist in the electrical power source will be avoided until the power source settles to its normal operating state. However, None of these devices coordinate the disconnect of the transducer and the remote turnoff control of the transducer&#39;s power source. 
     SUMMARY OF THE INVENTION 
     The current invention coordinates the electronic switching of transducers to their power sources in accordance with a remote turn on control signal and coordinates this action with the application of power to power sources and signal processing circuitry via remote turn on provisions. 
     In addition, the current invention senses and responds to the signal and rapidly connects and disconnects the transducer when the signal level drops to a level too low to be of any good effect. This action is accomplished in a novel manner that improves upon the manner implemented by prior art. 
     The present invention also removes the effect of signal level on delays provided for suppressing undesirable changes in response in a manner which improves upon the prior art. 
    
    
     BRIEF DESCRIPTIONS OF THE DRAWINGS 
     FIG. 1 is a block diagram of an exemplary audio reproduction system which includes the noise gate controller circuitry of the present invention; 
     FIG. 2 is a block diagram of the noise gate controller circuitry of the present invention; 
     FIG. 3 is an electronic schematic diagram of the gain stage, variable gain stage and second gain stage circuitry shown in FIG. 2; 
     FIG. 4 is an electronic schematic diagram of the attenuating circuitry included in the noise gate slave circuitry of FIG. 1; and 
     FIG. 5 is a timing diagram for the noise gate controller circuitry. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     FIG. 1 illustrates the utilization of the noise gate control circuitry of the present invention in a typical audio reproduction system. The input signals are representative of audio signals which are sensed by the noise gate controller 1 and user volume control 2. Signal Processor 3 and Amplifier 4 determine the electrical power excitation for the transducer of speakers 6. Noise gate slave switch 5 which is part of the noise gate control circuitry of the present invention, determines the application or removal of power from amplifier 4 to speakers 6. Noise gate controller 1 also determines power controls for signal processor(s) 3 and amplifier(s) 4 in such a manner as to eliminate turn on and turn off effects in speaker 6. Controller 1 also determines power control signals for amplifiers that may not use switch 5 to interrupt the flow of power between a given amplifier and speaker. 
     With reference to FIG. 2, the input signal is applied to an optional gain stage 7 to amplify the signal to enable circuit operation about low signal levels. Variable gain stage 8 provides for the adjustment of the sensitivity of the overall device in order to provide for higher signal levels for subsequent processing circuits. Gain stage 9 provides additional gain and the opportunity for additional response filter tailoring. By employing three separate gain stages higher overall attainable gain with lower gain inaccuracies and drifts are possible. 
     With reference to FIG. 3 the circuitry for each gain stage is shown. Source left and right signals are applied across resistor 21 and resistor 30. Resistor 22 and Capacitor 23 combine to filter high frequency noise interference in the left channel in the same manner as resistor 31 and capacitor 32 do in the right channel. Operational amplifier 25 and 34 buffer said inputs in order to provide accurate, low source impedance signals that can be accurately summed by the summing amplifier consisting of operational amplifier 42 and resistors 26,35,40,29,38 and 45. It is noted that resistors 29 and 38 are not separately required except that in this instance the inputs are isolated to avoid the possibility of interference currents between there ground connections which may be caused, for example, by the impression of stray magnetic fields upon the input cable conductors. As such two resistors, 29 and 38, sum the ground voltage reference potentials of the two inputs simultaneously. 
     In addition, operational amplifier 42 in conjunction with diodes 43 limits the sum calculation to about 0.6 volts peak-to-peak to prevent excess drive and overload from affecting the response of succeeding gain stages. Potentiometer 46 provides for a gain adjustment in advance of the threshold detection circuitry. In this way the threshold detection electronics that follow need only work about a given fixed level. Hence timing errors and inaccuracies are also fixed to a large degree. Threshold of sensitivity, therefore, is inversely related to the position of the potentiometer. For example, potentiometer positions which are high will yield higher sensitivities, but effect the lowest thresholds of sensitivity. Conversely, potentiometer positions which are low will yield lower sensitivities, but effect the highest threshold of sensitivity. At the full minimum position it is impossible to trigger the circuits as no significant signal will pass on to the detector circuitry. 
     Continuing to refer to FIG. 3, operational amplifier 49 implements the first stage of the pre-compensation and AC gain circuit. Resistor 47 and capacitor 48 provide additional high frequency interference filtering while capacitors 50 and 54 with resistors 41 and 53 provide for high AC gain that is tailored to result in an overall responsivity that approximates the relative sensitivity versus frequency of the human ear at low listening levels. Diodes 52 limit the output voltage of the operational amplifier and so provide for rapid charging and discharging of capacitor 54 to make high speed recovery from overloads in that circuit possible. The combination of resistor 53 and capacitor 54 provide for an approximate 6 db per octave roll off that begins the optimum response tailoring at low frequencies. Resistor 51 and capacitor 50 provide for an approximate 6 db per Octave roll off at high frequencies to appropriately tailor the response for those frequencies. 
     Operational amplifier 65 similarly determines the required response tailing. Like the previous gain stage, resistor 61 and capacitor 62 provide additional 6 db per octave low cut filtering to achieve an approximate 12 db per octave low cut filtering in combination with the effects of the previous stage in order to simulate the more dramatic drop-off of human hearing sensitivity at low levels and low frequencies. Diodes 60 like diodes 52 provide voltage limiting characteristics that prevent capacitor 62 from charging out of the range in which the gain of the circuit operates optimally. 
     Referring to FIGS. 2 and 5, a signal detector 10 is employed which may consist of a simple diode detector or a modern operational amplifier based precision full wave rectifier. Such a circuit will convert an input signal such as is illustrated by waveform 71 in FIG. 5 to a waveform of the type indicated by waveform 74. The result of the detector circuit is a uni-polar signal which is an indicator of the level of the signal. 
     As shown in waveform 71 of FIG. 5, exemplary signal waveform 71 includes a large interference spike 72 not unlike that frequently encountered in practice, and which has an undesirable effect on the system performance under ordinary circumstances. To avoid this undesirable effect, the noise gate controller 1 of the present invention includes a non-linear impulse filter 11. Non-linear impulse filter 11 has the characteristic of responding with a maximum rate of rise while the fall rate is not as severely limited. As a result, the effect of the limited rate of rise can be seen on the output of the filter circuit as shown in waveform 75 of FIG. 5. The impulse filter 11 may be implemented with a diode driving a shunt capacitor whereas the diode only conducts when the signal is falling. A resistor across the diode would determine the rise rate while the fall rate is limited only upon the diode characteristic and the impedance of the source driving it. Other implementations with greater precision such as those using operational amplifiers to eliminate the inherent voltage drop of the diode are known in the art. 
     The output of filter 11 is converted to a digital form by analog comparator 12, shown in FIG. 2. As shown in waveform 76 of FIG. 5, the comparator output responds to the level of impulse filter output 75 whenever this input crosses the comparator threshold as indicated by the dotted line. 
     After conversion to a digital pulse format, pulse compressor 13 counts the comparator output pulses and is employed to provide yet further precautions against false triggers due to interference noises such as spike 72 shown in waveform 71. Considering that the desired signal consists of many repetitions, pulse suppressor counts these repetitions and ignores a given count since the expiration of the last trigger event. It may be noted that the comparator 12 signal output illustrate by wave form reference 76 is stripped of the first pulse in each of the two pulse trains as shown by reference 77. This circuit may be implemented in a multitude of fashions. Since it is a completely digital circuit it may be implemented by digital counters, although the preferred method would be an analog pulse to staircase generator followed by a comparator with an adjustable threshold voltage to permit the simple implementation of a potentiometer controllable count. It may be noted that the count performed by this circuit must be reset in preparation for the next pulse inhibition cycle. As a result, the diagram of FIG. 2 shows the output of the hysteresis comparator 14 providing for this reset function. 
     In addition, the preferred embodiment includes a low pass filter 13A which provides for a degree of delay and additional noise immunity. The output of low pass filter 13A is transferred to hysteresis comparator 14 as shown in FIG. 2. Preferably, comparator 14 converts the undulations of the low pass filter to digital form in a way that results in a faster rise than fall in order to result in longer on state sustains as compared to off state sustain times when subject to normal signals. This is simply accomplished with hysteresis thresholds that are a small fraction of the voltage serving the low pass filter. When operating in the vicinity of smaller voltages such comparator circuits have much faster rise times than fall times. Since the fall times will relate to the on state sustain times, and the rise times relate to the off state sustain time in this instance, it should be apparent to those skilled in the art that such an effect is easily accomplished. 
     FIG. 5 illustrates the output waveform 78 of low pass filter 13A and the pair of doted lines illustrate the hysteresis threshold pairs which yield the waveform shown in waveform 79. It should be noted that a desirable reduction in the transition of states has occurred in the digital signal thus arrived at. 
     At this point the signal is processed with digital time domain analog circuits 15 and 16 that first extend the on time of the signal by a specified amount via on sustain circuit 15. In this way unwanted spurious transitions such as those illustrated at the beginning of the waveform 79 of FIG. 5 are effectively eliminated at the cost of extending the on state time of the control. It should be noted that some unwanted spurious transitions still occur at 82 in the example illustration of FIG. 5. 
     Following the extension of the on state time, a similar circuit is employed to provide for a programmed extension to the off state time via off sustain circuit 16. Off sustain circuit 16 is provided to prevent the activation of the attenuation circuit until a fixed time has elapsed whereby the preceding circuits are able to arrive at more a steady state before their indications affect the state of the attenuating circuit. The advantages of these fixed sustain times may be thought of as time guard bands that permit the previous circuits to settle. Alternatively, these times may be regarded as time domain hysteresis that prevents spurious state transitions within given minimum time intervals. In this way the transducers could never be gated faster than would be tolerated. 
     In addition to processing the signals in a manner described above, the present invention may include logic which coordinates operation of the noise gate control circuitry with the application and removal of system power so as to minimize any undesirable consequences to the transducer and the physical environment it is directed to, and is provided in turn on/turn off logic 17. Accordingly, the present invention includes turn on/turn off logic 17 for determining the power states of the transducer amplifier 4, controlled by the attenuating gate 90 as it relates to any signal processors or sources and the action of the noise gate controller. By sustaining the application of power to the source, and hence removing the power only after the attenuating gate has closed, power removal interference will not affect the transducer. Likewise, by delaying the release of the attenuating gate until after the related circuitry has settled down, the turn on/turn off logic 17 will do the same, i.e. will not affect the transducer, for any power on interference. Such an arrangement of timing provides a guard band of time separating the effects of the application and removal of power from the active intervals of the transducer. These timing relationships are shown at 84 and 85 in FIG. 5, where the immediate turn on and delayed turn off logic is illustrated, and at 86 and 87 where the delayed turn on logic is generated, which signal is logically ANDed with the results of the off sustain output waveform 81 to yield the final gate control output shown at 88 and 89 of FIG. 5. 
     Driver circuits 18, 19 and 20 of FIG. 2 provide for gate control signals, power control for signal path devices with gates, and power control for signal paths that are not equipped with attenuating gates, respectively. The gate control driver circuit 18 of FIG. 2 drives the light emitting diode 67 of the attenauting gate 90, shown in FIG. 4. Light emitting diode 67 thus eliminates diodes 68 which generate the voltage and energy necessary to fully engage power, MOSFETs 69 and 70. In this way power currents can flow freely between the transducer and the source bi-directionally when the subject attenuating gate 90 is engaged. Other methods of higher speed may be adopted which provide for greater power to more rapidly charge and discharge the MOSFET gates as may occur to those skilled in the art. 
     It will be understood that various modifications can be made to the embodiments of the present invention herein disclosed without departing from the spirit and scope thereof. Therefore, the above description should not be construed as limiting the invention but merely as exemplifications of preferred embodiments thereof. Those skilled in the art will envision other modifications within the scope and spirit of the present invention as defined by the claims appended hereto.