Abstract:
An active transconductance is provided by converting an input voltage into a current, and providing the current to a node which is maintained at a generally fixed voltage. Current is mirrored from the fixed voltage node to an output node. Such an active transconductance circuit can meet conventional performance specifications, but at a lower supply current, and/or with lower circuit complexity, and/or with a lower circuit area requirement.

Description:
TECHNICAL FIELD OF THE INVENTION 
     The invention relates generally to active electronic circuits and, more particularly, to active transconductance circuits. 
     BACKGROUND OF THE INVENTION 
     Active transconductance circuits are generally characterized by high linearity and high output impedance. These characteristics are desirable at the input of an active RC filter, for example, a radio receiver channel-select filter. 
       FIG. 1  diagrammatically illustrates an example of a conventional active RC low-pass filter. If the operational amplifier A is assumed to be ideal (infinite DC gain and bandwidth), then the filter response is entirely determined by the resistor ratio (α) and the RC product. However, a non-ideal operational amplifier will affect the filter response. For a variable gain implementation, the filter response even depends upon the gain setting. 
       FIG. 2  diagrammatically illustrates a conventional solution which reduces the non-ideal operational amplifier&#39;s impact on the filter response. In  FIG. 2 , the input resistance R/a of  FIG. 1  is replaced by an active input transconductance a/R. The high output impedance of the transconductance eliminates the influence of the resistor R/a on the filter response (see also  FIG. 1 ). Although the non-ideal operational amplifier does affect the filter response in  FIG. 2 , this is independent of the gain setting of the filter, and the effect is less significant than in the  FIG. 1  filter. 
     Ever increasing cost reductions in integrated circuits require continued migration toward ever smaller sized processes, which operate at ever lower supply voltages. These low supply voltages mean that the circuits must be able to operate with limited headroom. Voltage clipping due to limited headroom is a primary cause of harmonic distortion in active filters. 
     Although conventional active transconductances have the aforementioned advantages of high linearity and high output impedance, they nevertheless tend to produce harmonic distortion in active filters, to the extent that their behavior is non-linear and the available headroom is limited. Conventional active transconductances are also generally rather complex circuits whose performance relies on good component matching. 
     SUMMARY OF THE INVENTION 
     It is desirable in view of the foregoing to provide an active transconductance circuit that can meet conventional performance specifications, but at a lower supply current, and/or with lower circuit complexity, and/or with a lower circuit area requirement. 
     To address the above-discussed deficiencies of the prior art, it is a primary object of the present invention to provide an active transconductance circuit which, in exemplary embodiments, converts an input voltage into a current, and provides the current to a node that is maintained at a generally fixed voltage. Current is mirrored from the fixed voltage node to an output node. Such an active transconductance circuit is highly linear, has a high output impedance, requires relatively low supply current, relatively low circuit complexity and a relatively small amount of circuit area. Before undertaking the DETAILED DESCRIPTION OF THE INVENTION below, it may be advantageous to set forth definitions of certain words and phrases used throughout this patent document: the terms “include” and “comprise,” as well as derivatives thereof, mean inclusion without limitation; the term “or,” is inclusive, meaning and/or; the phrases “associated with” and “associated therewith,” as well as derivatives thereof, may mean to include, be included within, interconnect with, contain, be contained within, connect to or with, couple to or with, be communicable with, cooperate with, interleave, juxtapose, be proximate to, be bound to or with, have, have a property of, or the like; and the term “controller” means any device, system or part thereof that controls at least one operation. A controller may be implemented in hardware, firmware or software, or some combination of at least two of the same. It should be noted that the functionality associated with a controller may be centralized or distributed, whether locally or remotely. Definitions for certain words and phrases are provided throughout this patent document, those of ordinary skill in the art should understand that in many, if not most instances, such definitions apply to prior, as well as future uses of such defined words and phrases. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       For a more complete understanding of the present invention and its advantages, reference is now made to the following description taken in conjunction with the accompanying drawings, in which like reference numerals represent like parts: 
         FIG. 1  diagrammatically illustrates an active RC low-pass filter according to the prior art; 
         FIG. 2  diagrammatically illustrates an active RC low-pass filter with a transconductor according to the prior art; 
         FIG. 3  is a block diagram of a transconductor according to exemplary embodiments of the invention; 
         FIG. 4  diagrammatically illustrates an active RC low-pass filter including a transconductance according to exemplary embodiments of the invention; 
         FIG. 5  diagrammatically illustrates a fully-differential transconductance circuit according to exemplary embodiments of the invention; 
         FIG. 6  diagrammatically illustrates a portion of  FIG. 5  in more detail according to exemplary embodiments of the invention; and 
         FIG. 7  diagrammatically illustrates a further portion of  FIG. 5  in more detail according to exemplary embodiments of the invention. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       FIGS. 1 through 7 , discussed herein, and the various embodiments used to describe the principles of the present invention in this patent document are by way of illustration only and should not be construed in any way to limit the scope of the invention. Those skilled in the art will understand that the principles of the present invention may be implemented in any suitably arranged processing system. 
       FIG. 3  diagrammatically illustrates a transconductor according to exemplary embodiments of the invention. A resistor  34  is applied directly at the voltage input Vin. This resistor  34 , having a resistance value of R/α (see also  FIGS. 1 and 2 ) functions as a linear voltage-to-current converter. A voltage control circuit  31  is connected to the resistor  34  at a terminal  35  thereof opposite the voltage input terminal. The voltage control circuit  31  operates to maintain the resistor terminal  35  generally fixed at a common-mode voltage, which can be considered as a virtual ground. The resulting current, designated generally at  36 , is essentially entirely determined by the resistor  34 . A current mirror  32  mirrors the current at  36  to the output  33 . The transconductance circuit  30  has a high output impedance, which can result in an accurate filter response that is independent of the filter gain setting. The voltage control circuit  31  maintains the node  35  generally fixed at a common-mode voltage based on a common-mode reference voltage input, Vcm. 
       FIG. 4  diagrammatically illustrates an active RC low-pass filter according to exemplary embodiments of the invention. The filter of  FIG. 4  is generally similar to the filter of  FIG. 2 , but uses the transconductance circuit  30  of  FIG. 3 , with the output  33  connected to the inverting input of the operational amplifier A. A resistor R and capacitor C are connected in parallel between the inverting input and the output of the operational amplifier A. The non-inverting input of the operational amplifier A is connected to ground potential in the example of  FIG. 4 . The linear voltage-to-current conversion operation performed by the resistor R/a (see also  FIG. 3 ) provides the filter of  FIG. 4  with high linearity. The high output impedance provided by the transconductance circuit  30  provides a highly accurate frequency response. 
       FIG. 5  diagrammatically illustrates a fully-differential transconductance circuit according to exemplary embodiments of the invention. In  FIG. 5 , the voltage control circuit  31  of  FIG. 3  is implemented by a common-mode feedback (CMFB) circuit  51  together with transistors M 1   a  and M 2   a . For current mirror operations, current sources at  52  and  53  each provide a generally constant current I 1 . These current sources  52  and  53  are connected to the respective drains of the transistors M 1   a  and M 1   b . Also provided are current mirror transistor pairs M 2   a ,M 3   a  and M 2   b ,M 3   b . The drain of transistor M 3   a  forms the positive component Ioutp of the differential output, and the drain of transistor M 3   b  forms the negative component Ioutn of the differential output. The sources of transistors M 1   a  and M 1   b  are connected to the drains of transistors M 2   a  and M 2   b , respectively. 
     For current mirror operations, current sources at  54  and  55  each provide a generally constant current I 2  to the drain of a respectively connected one of the transistors M 3   a  and M 3   b . Bias voltages Vbiasp and Vcascp control the current sources  52 - 55  according to well-known conventional techniques. 
     Respective source-drain connections at M 1   a ,M 2   a  and M 1   b ,M 2   b  define fixed voltage nodes n 1   a  and n 1   b , respectively, where the differential input currents through resistors  34  are added to the constant current I 1 . The resulting differential current flows through transistors M 2   a  and M 2   b , and is mirrored by the output transistors M 3   a  and M 3   b  to produce the differential output current at Ioutp and Ioutn. In some embodiments, the transistors within each of the three transistor pairs M 1   a ,M 1   b , and M 2   a ,M 2   b , and M 3   a ,M 3   b  are identical in design. In some embodiments, all transistors are MOS transistors. 
     The voltage-to-current conversion performed by the resistors  34  on the differential input voltages Vinn and Vinp is a linear operation. This results in a highly-linear transconductor circuit. If the current sources  52 - 55  are assumed to be ideal, then the transconductance of the circuit is determined by the resistance R/a of the resistors  34 , and the current ratio N between (W/L) 3  and (W/L) 2  as follows: 
                   G   =             (     W   L     )     3     /       (     W   L     )     2             2   ⁢   R     α     +       2   ⁢       g     o   ⁢           ⁢   1       ⁡     (     1   +       g     o   ⁢           ⁢   2       ⁢   R       )             g     m   ⁢           ⁢   2       ⁡     (       g     m   ⁢           ⁢   1       +     g     o   ⁢           ⁢   1         )             ≈       α   ⁢           ⁢   N       2   ⁢   R                 (   1   )               
where (W/L) 3  represents the width-to-length ratio of the transistors of transistor pair M 3   a ,M 3   b , and (W/L) 2  represents the width-to-length ratio of the transistors of transistor pair M 2   a ,M 2   b.    
     The output current Ioutn, Ioutp of the transconductance circuit of  FIG. 5  needs to be large enough to obtain the required maximum output voltage at the filter output (see also  FIG. 4 ). Basically, any required output current is achievable by simply scaling the circuit. 
     The common-mode feedback (CMFB) circuit  51  senses a common-mode voltage associated with nodes n 1   a  and n 1   b , compares the sensed voltage to a reference voltage, and controls the gates of transistors M 1   a  and M 1   b  to keep nodes n 1   a  and nib at a generally fixed voltage. 
     The voltage sources Vlvs connected to the drains of transistors M 1   a  and M 1   b , and further connected to the gates of transistors M 2   a  and M 2   b , represent level shifters  56  which keep transistors M 1   a  and M 1   b  in saturation. 
       FIG. 6  diagrammatically illustrates the CMFB circuit  51  of  FIG. 5  in more detail according to exemplary embodiments of the invention. In  FIG. 6 , transistors M 45  and M 46  are biased appropriately by signals  63  and  64  to operate in the linear region, and to therefore act as a resistive voltage divider. Accordingly, the common-mode voltage level associated with nodes n 1   a  and n 1   b  is present at the center tap  65  of the resistive voltage divider. The remainder of the circuitry in  FIG. 6 , consisting of transistors M 28 , M 47 , M 48 , M 49 , M 52 , M 53  and M 54 , together with capacitor C 2 , all connected as shown, constitutes an operational amplifier circuit which compares the common-mode voltage level at center tap  65  to the reference voltage Vcm. The output  61  of the operational amplifier provides an output signal that drives the gates of transistors M 1   a  and M 1   b  of  FIG. 5 . The bias voltage  62  controls transistors M 28  and M 47  according to well-known conventional techniques. In some embodiments, C 2 =0.5 pF. 
     Other specific implementations of CMFB circuit  51  can also be used, without constraining performance. 
       FIG. 7  diagrammatically illustrates in more detail exemplary embodiments of the level shifters  56  of  FIG. 5 . As shown in  FIG. 7 , each level shifter  56  includes transistors M 38 , M 39  and M 42  connected in a source follower configuration. The source followers shift the DC levels as illustrated in  FIG. 5 . The level shifters  56  also include a capacitor C 0  connected across the terminals  71  and  72  thereof. In some embodiments, C 0 =4.5 pF. Bias voltages  73  and  74  control transistors M 38  and M 39  according to well-known conventional techniques. 
     Other specific implementations of the level shifters  56  can also be used, without constraining performance. 
     A mismatch between resistors  34  in the embodiments of  FIGS. 5-7  can cause an imbalance between the current through those resistors, thereby causing even-order harmonics that in turn are copied to the outputs. A mismatch between transistors M 1   a  and M 1   b  causes an offset between the virtual grounds at n 1   a  and n 1   b , thereby causing the same effect as a mismatch between the resistors  34 . Because the aforementioned mismatches essentially affect the virtual ground references, and not a relatively large signal voltage, the impact of such mismatches can be expected to be less significant than in conventional transconductor circuits. 
     Simulations have shown that, with respect to operational specifications such as linearity, filter accuracy over gain range, robustness against component mismatches, and harmonic distortion, transconductance circuits according to the embodiments of  FIGS. 3-7  can provide the same operational specifications as conventional transconductor circuits, but with lower circuit complexity, lower circuit area requirements, and lower supply current requirements. 
     As described above, exemplary embodiments of the invention provide a linear, low-complexity transconductor circuit which is suitable for use in accurate variable-gain filter applications at low supply voltages. Transconductor circuits according to the invention combine high linearity with relatively low circuit complexity and relatively low supply current requirements. 
     Although the present invention has been described with an exemplary embodiment, various changes and modifications may be suggested to one skilled in the art. It is intended that the present invention encompass such changes and modifications as fall within the scope of the appended claims.