PATENT DOCUMENT

Publication Number: US-10284089-B2
Application Number: US-201715452411-A
Country: US
Kind Code: B2

Title: Integrated bi-directional driver with modulated signals

Abstract:
According to some embodiments, a bi-directional converter is configured to operate in a boost mode or a buck mode. The bi-directional converter includes a hysteresis control unit that includes a comparator that can be configured to determine whether the reference voltage has a positive slope or a negative slope in conjunction with the bi-directional converter operating in boost mode or buck mode. In addition, the comparator is configured to compare a fractional load voltage to a reference voltage so that the output load voltage corresponds to the waveform shape of the reference voltage.

Claims:
What is claimed is: 
     
       1. A bi-directional actuator driver circuit comprising:
 a first switch; 
 a second switch; 
 an output node that is configured to be coupled to an actuator; 
 a first input node that is configured to be coupled to a power supply; 
 a second input node that is configured to provide, to a hysteresis control unit, a fractional voltage of a load voltage, wherein the hysteresis control unit is configured to: i) compare the fractional voltage to a reference voltage from a waveform generator configured to generate the shape of the waveform of the reference voltage and corresponding actuator output and ii) generate an output value based on the comparison; and 
 a control logic module being configured: i) to provide a first control signal to the first switch, ii) to provide a second control signal to the second switch, and iii) to cause the bi-directional actuator driver to:
 operate in a boost mode when a slope of the reference voltage is positive and otherwise operate in a buck mode so that an output voltage at the output node corresponds to the waveform shape of the reference voltage. 
 
 
     
     
       2. The bi-directional actuator driver circuit of  claim 1 , wherein during the boost mode, the first switch is toggled via the first control signal that is actively switching and the second switch is open via the second control signal that is in an un-asserted state. 
     
     
       3. The bi-directional actuator driver circuit of  claim 1 , wherein during the buck mode, the second switch is toggled via the second control signal that is actively switching and the first switch is open via the first control signal that is in an un-asserted state. 
     
     
       4. The bi-directional actuator driver circuit of  claim 1 , wherein the hysteresis control unit configured to provide a hysteresis amount to prevent excessive switching. 
     
     
       5. The bi-directional actuator driver circuit of  claim 1 , wherein the reference voltage is a sinusoidal waveform or a trapezoidal waveform. 
     
     
       6. The bi-directional actuator driver circuit of  claim 1 , wherein the power supply is a battery. 
     
     
       7. The bi-directional actuator driver circuit of  claim 1 , wherein the reference voltage is characterized by a reference voltage frequency, and the reference voltage frequency is less than a switching frequency of the first control signal. 
     
     
       8. The bi-directional actuator driver circuit of  claim 1 , wherein the fractional voltage of the load voltage is determined by a voltage divider that is coupled to the second input node. 
     
     
       9. A portable electronic device comprising:
 a power supply; 
 a load; and 
 a bi-directional actuator driver comprising:
 a first switch, 
 a second switch, 
 an output node coupled to the load, 
 a first input node coupled to the power supply, 
 a second input node that is configured to provide, to a hysteresis control unit, a fractional voltage of a load voltage, wherein the hysteresis control unit is configured to: i) compare the fractional voltage to a reference voltage from a waveform generator configured to generate the shape of the waveform of the reference voltage and corresponding actuator output and ii) generate an output value based on the comparison, and 
 a control logic module being configured: i) to provide a first control signal to the first switch, ii) to provide a second control signal to the second switch, and iii) to cause the bi-directional actuator driver to:
 operate in a boost mode when a slope of the reference voltage is positive and otherwise operate in a buck mode so that an output voltage at the output node corresponds to the waveform shape of the reference voltage. 
 
 
 
     
     
       10. The portable electronic device of  claim 9 , wherein during the boost mode, the first switch is toggled via the first control signal that is actively switching and the second switch is open via the second control signal that is in an un-asserted state. 
     
     
       11. The portable electronic device of  claim 9 , wherein during the buck mode, the second switch is toggled via the second control signal that is actively switching and the first switch is open via the first control signal that is in an un-asserted state. 
     
     
       12. The portable electronic device of  claim 9 , wherein the hysteresis control unit configured to provide a hysteresis amount to prevent excessive switching. 
     
     
       13. The portable electronic device of  claim 9 , wherein the reference voltage is a sinusoidal waveform or a trapezoidal waveform. 
     
     
       14. The portable electronic device of  claim 9 , wherein the power supply is a battery. 
     
     
       15. The portable electronic device of  claim 9 , wherein the reference voltage is characterized by a reference voltage frequency, and the reference voltage frequency is less than a switching frequency of the first control signal. 
     
     
       16. The portable electronic device of  claim 9 , wherein the fractional voltage of the load voltage is determined by a voltage divider that is coupled to the second input node. 
     
     
       17. A method for switching a bi-directional actuator driver between a boost mode and a buck mode, the method comprising:
 receiving a fractional voltage of a load voltage; 
 comparing the fractional voltage to a reference voltage from a waveform generator configured to generate the shape of the waveform of the reference voltage and corresponding actuator output; 
 generating an output value based on the comparison; 
 determining a slope of the reference voltage; 
 determining an operational mode based on the slope, wherein the operational mode is a boost mode when the slope is positive and the operational mode is a buck mode otherwise; 
 providing a first control signal based on the output value to a first switch during the boost mode; and 
 providing a second control signal based on the output value to a second switch during the buck mode; 
 wherein the actuator driver generates an output voltage corresponding to the waveform shape of the reference voltage. 
 
     
     
       18. The method of  claim 17 , wherein during the boost mode, the first switch is toggled via the first control signal that is actively switching and the second switch is open via the second control signal that is in an un-asserted state. 
     
     
       19. The method of  claim 17 , wherein during the buck mode, the second switch is toggled via the second control signal that is actively switching and the first switch is open via the first control signal that is in an un-asserted state. 
     
     
       20. The method of  claim 17 , wherein the reference voltage is a sinusoidal waveform or a trapezoidal waveform.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     The present application claims the benefit of U.S. Provisional Application No. 62/399,050, entitled “INTEGRATED BI-DIRECTIONAL DRIVER WITH MODULATED SIGNALS” filed Sep. 23, 2016, the content of which is incorporated herein by reference in its entirety for all purposes. 
    
    
     FIELD 
     The described embodiments are directed towards a bi-directional driver and techniques for controlling the bi-directional driver to operate in a boost mode and a buck mode. 
     BACKGROUND 
     Portable electronic devices may include an actuator requiring a high voltage in order to drive the actuator. Usually, the high voltage may be generated from a single cell battery having a low starting voltage. Conventional solutions to generate the high voltage from a low starting voltage may utilize a switching boost converter to generate a voltage rail at a level higher than a peak voltage to be supplied to the load. However, these conventional solutions can require high voltage capacitors to generate the necessary high voltage. For example, conventional switching boost converters may not be able to sustain the full amount of power required by the load over a prolonged period of time due to the small battery size of portable electronic devices. In addition, the energy that is provided to the load can be offset with a large amount of voltage drop. In order to efficiently drive the load for a prolonged period of time, power management circuits may need to include large capacitors and/or generate a large input current which can be impractical given the small form factor of many of the current portable electronic devices. Other conventional solutions include a class AB or class D amplifier to generate the necessary waveform. However, the energy stored by a capacitive load is often dissipated by the amplifier so that no energy recovery can be accomplished. 
     SUMMARY 
     This paper describes various embodiments related to a power management system including a bi-directional driver. Techniques for switching the bi-directional driver between a boost mode and a buck mode are described herein. 
     A bi-directional converter is provided herein including an inductor coupled to two switches. The bi-directional converter is positioned between a power supply and a load. The switches are in parallel with diodes. A first switch is in shunt configuration, and the second switch is in series between the inductor and a load. Control circuitry compares a reference signal with a sensed version of the output voltage on the load. Based on the comparison, pulse trains are sent in a bursty fashion to one or the other switch. In a boost mode, the first (configured as a shunt switch) switch allows current to flow through the inductor. When the first switch opens, the current flows through a diode, bypassing the second switch, and into the load, resulting in a boosted output voltage. In a buck mode, the second switch allows a current to flow from the load through the inductor to the power supply. In addition, when the second switch opens, the collapsing magnetic field in the inductor provides an additional current flow into the power supply through the diode bypassing the first switch. 
     Provided herein is a method for switching a bi-directional converter between a boost mode and a buck mode. The method includes receiving a fractional voltage of a load voltage, comparing the fractional voltage to a reference voltage, generating an output value based on the comparison, determining a slope of the reference voltage, and determining an operational mode based on the slope, wherein the operational mode is a boost mode when the slope is positive and otherwise the operational mode is a buck mode. The method provides a first control signal based on the output value to a first switch during the boost mode and provides a second control signal based on the output value to a second switch during the buck mode. 
     The described embodiments may be better understood by reference to the following description and the accompanying drawings. Additionally, advantages of the described embodiments may be better understood by reference to the following description and accompanying drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The disclosure will be readily understood by the following detailed description in conjunction with the accompanying drawings, wherein like reference numerals designate like structural elements, and in which: 
         FIG. 1  illustrates an exemplary electronic device that includes a bi-directional converter, in accordance with some embodiments. 
         FIG. 2A  illustrates a circuit diagram of a bi-directional converter configured to re-store energy to a power supply, e.g., a battery, using two switches, in accordance with some embodiments. 
         FIG. 2B  illustrates an exemplary hysteresis control unit, according to some embodiments. 
         FIG. 2C  illustrates an exemplary output voltage waveform and time intervals in which boost and buck modes are invoked by the circuit of  FIG. 2A , according to some embodiments. 
         FIG. 3  illustrates exemplary relationships between a reference waveform and an output waveform produced based on operation of the bi-directional converter, where the output waveform is represented by a sense waveform, in accordance with some embodiments. 
         FIG. 4  illustrates states of the two switches and circuit currents during operation of the circuit of  FIG. 2A  in boost mode, according to some embodiments. 
         FIG. 5  illustrates states of the two switches and circuit currents during operation of the circuit of  FIG. 2A  in buck mode, according to some embodiments. 
         FIG. 6A  illustrates exemplary components of the inductor current of  FIG. 2A  in relation to a switching waveform applied to the first switch in accordance with  FIG. 4  (boost mode), according to some embodiments. 
         FIG. 6B  illustrates exemplary components of the inductor current of  FIG. 2A  in relation to a switching waveform applied to the second switch in accordance with  FIG. 5  (buck mode), according to some embodiments. 
         FIG. 7  illustrates a flowchart for operating the bi-directional converter, in accordance with some embodiments. 
     
    
    
     Those skilled in the art will appreciate and understand that, according to common practice, various features of the drawings discussed below are not necessarily drawn to scale, and that dimensions of various features and elements of the drawings may be expanded or reduced to more clearly illustrate the embodiments of the present invention described herein. 
     DETAILED DESCRIPTION 
     The following disclosure describes various embodiments of a bi-directional converter and techniques for operating the bi-directional converter. Certain details are set forth in the following description and figures to provide a thorough understanding of various embodiments of the present technology. Moreover, various features, structures, and/or characteristics of the present technology can be combined in other suitable structures and environments. In other instances, well-known structures, materials, operations, and/or systems are not shown or described in detail in the following disclosure to avoid unnecessarily obscuring the description of the various embodiments of the technology. Those of ordinary skill in the art will recognize, however, that the present technology can be practiced without one or more of the details set forth herein, or with other structures, methods, components, and so forth. 
     A bi-directional converter is set forth herein which includes a power management function. That is, the bi-directional converter boosts current originating at a source battery, where the boost current is needed to produce a high voltage at a load terminal. The bi-directional converter also recovers energy from the load to be restored to the battery, while satisfying load waveform requirements. 
     By combining the buck converter and boost converter into a bi-directional converter, the bi-directional converter significantly reduces the number of switching components. In this instance, excessive switching delays normally associated with a power management circuit having a separate buck converter and a separate boost converter operating in buck mode and boost mode are significantly reduced. In addition, the disclosed bi-directional converter is configured to deliver a boost current to a load during a boost mode and provide a charging current to the battery in a buck mode. In this way, the battery can efficiently recover energy that would otherwise be lost. By reducing the number of electronic components in the bi-directional converter, a reduced topology in terms of circuit board area is achieved. The bi-directional converter, which consumes little space, is useful in a portable electronic device. 
     The system and methods described herein can be used to perform power management functions for computers, portable electronic devices, wearable electronic devices, server devices, computer network storage devices, and general electronic devices, such as those manufactured by Apple Inc., based in Cupertino, Calif. 
     These and other embodiments are discussed below with reference to  FIGS. 1-7 . The detailed description given herein with respect to these figures is for explanatory purposes only and should not be construed as limiting. The numerical values used in this discussion are purely illustrative and are for purposes of examples only, and a wide range of values could be used in conjunction with the power management functions provided herein. 
     System 
       FIG. 1  illustrates a block diagram of an electronic device  100 , in accordance with some embodiments. Electronic device  100  includes a bi-directional converter  110  that is coupled to a power supply (e.g., battery)  120 . The bi-directional converter  110  can be configured to receive an input battery voltage from the power supply  120 . In some embodiments, processor  192  controls the functions of the bi-directional converter  110 . In some embodiments, logic internal to the bi-directional converter  110  controls the functions of the bi-directional converter  110 . The functions of the components shown in  FIG. 1  can be implemented by a combination of hardware and/or software. The electronic device  100  can optionally include one or more non-transitory computer-readable storage mediums or memory  190  for storing instructions related to executing the functions of the control logic that can be executed by one or more processors (e.g., processor  192 ). In some embodiments, the bi-directional converter  110  is electrically coupled to a load  130 . In some examples, the load  130  is an electromechanical actuator, a piezoelectric actuator, a sensor, or the like. 
     The load  130  can be characterized as having a small capacitance value, e.g., between about 20 nF to about 300 nF. In some embodiments, the load  130  can be characterized as operating with an alternating current (A/C) signal with a frequency range between about 5 KHz to about 200 KHz. In some examples, the A/C signal of the voltage (i.e., load voltage) that is applied to the load  130  can be characterized as having a waveform that is sinusoidal, trapezoidal, and the like. In some embodiments, the waveform shape of the output load voltage/input load voltage corresponds to the waveform shape of the reference voltage. 
     The power supply  120  can refer to a rechargeable battery, such as a lithium-ion battery pack, nickel metal hydride battery pack, and the like. Lithium-ion batteries are widely used in portable electronic devices because of their high energy density, long cycle life and the absence of memory effects. In some examples, the battery can have a voltage between e.g., 3 V and 15 V (e.g., the battery pack may experience a range of voltages during operation). In some instances, the power supply  120  can refer to power adapters that provides an input battery voltage that is either less than the lowest battery pack voltage (e.g., 5 V when the battery voltage is between 6 V and 8.7 V) or greater than the highest battery pack voltage (e.g., 12 V or 15 V when the battery voltage is between 6 V and 8.7 V). For example, a range of the input battery voltage can be between e.g., 0.1 V to 20 V. More specifically, the range of the input battery voltage can be between, e.g., 5 V to 12 V. 
     In some embodiments, the components of the electronic device  100  can be electrically coupled by signal lines, links or buses  102 . While electrical communication has been used as an illustrative example, in general these connections may include electrical, optical, or electro-optical communication of signals and/or data. Furthermore, in the preceding embodiments, some components are shown directly connected to one another, while others are shown connected via intermediate components. In each instance the method of interconnection, or ‘coupling,’ establishes some desired communication between two or more circuit nodes, or terminals. Such coupling may often be accomplished using a number of circuit configurations, as will be understood by those of skill in the art (e.g., AC coupling and/or DC coupling may be used). 
     In some embodiments, functionality of a bi-directional converter, components, and the electronic device can be implemented in one or more: application-specific integrated circuits (ASICs), field-programmable gate arrays (FPGAs), and/or one or more digital signal processors (DSPs). Moreover, the circuits and components may be implemented using any combination of analog and/or digital circuitry, including: bipolar, PMOS and/or NMOS gates or transistors. Furthermore, signals in these embodiments may include digital signals that have approximately discrete values and/or analog signals that have continuous values. Additionally, components and circuits may be single-ended or differential, and power supplies may be unipolar or bipolar. 
     Bi-Directional Converter 
       FIG. 2A  illustrates exemplary details of the bi-directional converter  110 , including identified currents and voltages useful for explaining operating modes, according to some embodiments. Details of a hysteresis control unit  250  are provided in  FIG. 2B . An illustration of a resulting output waveform in a boost mode  281  and a buck mode  282  is provided in  FIG. 2C . 
     As illustrated in  FIG. 2A , an exemplary bi-directional converter  110  includes controller  290 , the hysteresis control unit  250 , a first switch S 1    222 , a second switch S 2    224 , and an inductor  236 . The controller  290  includes control logic  220 , a pulse width modulator (PWM) unit  291  and a waveform generator  292 . The bi-directional converter  110  is coupled to the power supply  120  at a node  261  and to the load  130  at a node  263 . The inductor  236  couples the node  261  to an internal node, node  262 . The switch S 1    222  couples the node  262  to ground and is in parallel with a diode D 1    215  which is oriented with its anode terminal at ground. The switch S 2    224  couples the node  263  to the node  262  and is in parallel with a diode D 2    226  which is oriented with its anode terminal at the node  262 . The load  130 , in some embodiments, may be modelled as including a capacitor C  271 . Generally, the load  130  may also be modelled as including a resistor in series with C  271  (not shown). 
     Hysteresis control unit  250  receives inputs of V SENSE    217  from a resistive divider that provides a fractional value (V SENSE    217 ) of load voltage, V OUT    218 . The resistive divider is made up of R  241  and R  242 . R  242  is in parallel with a capacitor C  243 , in some embodiments. In some embodiments, resistor R  241 , resistor R  242 , and capacitor C  271  function as a voltage divider and a low pass filter between V OUT    218  and V SENSE    217 . V SENSE    217  is the voltage at a node formed by terminals of R  241 , R  242 , and C  243 . The other terminal of R  241  is at the node  263 . The other terminal of R  242  is at ground. 
     As mentioned above, the bi-direction converter  110  includes a voltage divider (shown as R  241  and R  242 ) that is coupled to the load  130 . The voltage divider output may be referred to herein as a fractional load voltage. In some examples, the fractional load voltage is an amplified version (1/divider_value) of the reference voltage that is established by the control logic  220 . 
     In some examples, the voltage divider can be incorporated within the bi-directional converter  110 . In other examples, the voltage divider can be external to the bi-directional converter  110  and the fractional voltage, V SENSE    217 , is provided to the hysteresis control unit  250 . 
     The hysteresis control unit  250 , in some embodiments, includes a comparator  253  (see  FIG. 2B ). The comparator  253  receives as inputs the signal V REF    216  and the signal V SENSE    217 . In some embodiments, the signal V REF    216  is provided by the waveform generator  292  (connection not shown in  FIG. 2A ). In some embodiments, V REF    216  is provided at an input to the bi-directional converter  110 , denoted node  258  in  FIG. 2A . The output of the control logic  220  includes signals CTRL 1    235  and CTRL 2    234 . In some embodiments, CTRL 1    235  and CTRL 2    234  are output signals of the PWM unit  291 . The hysteresis control unit  250  is configured to use the comparator  253  output, V CMP    254 , to provide control signals indicated as CTRL HYS    237  to the control logic  220  to operate the bi-directional converter  110  in buck mode  282  when the slope (Δvoltage/Δtime) of the reference voltage is negative (slope ≤0) (see  FIG. 2C ). The comparator  253  exhibits hysteresis and the levels of hysteresis are configurable based on a control signal CTRL LEVEL    252  from the control logic  220 . Control logic  220  and switching logic  251  are aware of V REF    216  and V SENSE    217 . In some instances, switching logic  251  sets CTRL HYS    237  based on one or both of these. In this manner, the control logic  220  can be configured to generate the desired shape of the waveform of the output load voltage with an acceptable level of ripple. 
       FIG. 2C  introduces an illustration of the waveform VOUT  218 . VOUT  218  follows, by operation of the bi-directional converter  110 , the reference waveform VREF  216 , which is generally periodic. One period of VOUT  218  is illustrated in  FIG. 2C . VOUT  218  generally includes some ripple; the ripple is not shown in  FIG. 2C . The y-axis corresponds to the intensity of VOUT  218  and the x-axis corresponds to time, indicated as time  283 . From a time T 1   285  to a time T 2   286 , the slope of VOUT  218  is positive and the bi-directional converter  110  operates in the boost mode  281 . From the time T 2   286  to a time T 3   287 , the slope of VOUT  218  is zero or negative and the bi-directional converter  110  operates in the buck mode  282 . VOUT  218  is lower-bounded by the value VBAT  208 . VOUT  218  experiences a peak value indicated as VPEAK  284  which occurs at the time T 2   286 . In some examples, VOUT  218  has a range between, e.g., about 1 V to about 150 V. In some examples, the peak voltage VPEAK  284  is at least, e.g., 100 V. In some instances, the peak voltage VPEAK  284  is between, e.g., 100 V to 150 V. 
     The bi-directional converter  110  uses the current I L    203  through the inductor  236  to boost energy to the load  130  (boost mode  281 ) or take energy from the load  130  (buck mode  282 ) so that the waveform V OUT    218  imitates the shape (other than amplitude scaling) of the waveform V REF    216  within acceptable limits. 
     The comparator  253  is a central piece in influencing V OUT    218  to follow V REF    216 . The comparator  253  can be configured to compare V SENSE    217  to V REF    216  to determine the intermediate value V CMP    254 . In some examples, the reference voltage waveform V REF    216  can be established by the control logic  220 , as mentioned above. Based on the comparison between the fractional load voltage, V SENSE    217 , and the reference voltage V REF    216 , the comparator  253  can generate the intermediate value V CMP    254 . The signal CTRL HYS    237  from the switching logic  251  to the control logic  220  depends on whether the bi-directional converter  110  is in boost mode  281  or buck mode  282  and depends on the intermediate value V CMP    254 . The control logic  220  can then cause the PWM unit  291  to emit one or more control signals, e.g., CTRL 1    235  and CTRL 2    234 , which are then applied in the bi-directional converter  110  to the switches S 1    222  and S 2    224 . Depending on which of the one or more control signals e.g., CTRL 1    235  and CTRL 2    234 , are asserted, the bi-directional converter will operate in boost mode  281  or buck mode  282  with V OUT    218  following V REF    216 . Opportunistic energy recovery to the power supply  120  can occur during buck mode  282 . 
     The control logic  220  can be configured to change the type of control signals that are provided by the PWM unit  291  depending on whether the bi-directional converter  110  is operating in boost mode  281  or buck mode  282 . For example, the control signals generated by the control logic  220  can be based on intermediate value V CMP    254  and the slope of V OUT    218  or of V REF    216 . Slope refers to a change in voltage with time, e.g., Δvoltage/Δtime. The control logic  220  is configured to adjust the duty cycle of the waveform of the electrical pulses generated by the PWM unit  291 , as described in greater detail with reference to  FIG. 6A . The PWM unit  291  can be configured to adjust the current that is provided to the load  130  by switching between a low period and a high period of a pulse duty cycle. In some embodiments, the PWM unit  291  can adjust the duty cycle of the control signals generated in conjunction with boost mode and buck mode to define a minimum and maximum pulse duty cycle. The PWM unit  291  adjusts the duty cycle of the control signals so that the V OUT    218  waveform shape corresponds to the V REF    216  waveform shape. 
     In some embodiments, the hysteresis control unit  250  can be configured to regulate the toggling, or switching activity, of the switches S 1    222  and S 2    224  to cause the V OUT    218  waveform to correspond to the V REF    216  waveform shape. Sending a train of pulses from the PWM unit  291  to a switch is referred to herein as active switching. Holding a control line at logic low level is referred to herein as an un-asserted control state. The hysteresis control unit  250  can be configured to control an amount of voltage ripple as the bi-directional converter  110  operates in either boost mode  281  or buck mode  282 , independent of a load capacitance, reference waveform shape, and/or reference waveform frequency. 
     In some embodiments, the waveform generator  292  can be configured to generate the shape of the waveform of the reference voltage V REF    216  and thus of the load output voltage V OUT    218  (other than, for example, ripple). Exemplary shapes include a sinusoidal or trapezoidal waveform shape. 
     In some examples, the electronic device  100  can refer to a tablet computer, a smartphone, a touch-sensitive device, a stylus, an electronic accessory, a portable computer, a smart watch, a consumer-electronic device, a digital organizer, a cellular phone, a network appliance, a server. 
       FIG. 2A , described above, illustrates exemplary power management functions of an electronic device, in accordance with some embodiments. The power management functions are provided by the bi-directional converter  110 . In some embodiments, a bi-directional converter  110  can include any number of switches, e.g., four switches. In some examples, the bi-directional converter  110  may be referred to as a buck-boost circuit. In some examples, a bi-directional converter  110  can be characterized as a DC-to-DC converter. As described above, the bi-directional converter  110  can include control logic  220 , where the control logic  220  is configured to generate one or more control signals to toggle or adjust the switches S 1    222  and S 2    224  to operate the bi-directional converter  110  in boost mode  281  or buck mode  282 . In some examples, the control logic  220  can be configured to cause the bi-directional converter  110  to operate in a charge configuration (i.e., boost) to generate an input current from the power supply  120  to cause an output load voltage at the load  130 . Then, the control logic  220  can cause the bi-directional converter  110  to operate in a load discharge configuration (i.e., buck) to step down a load voltage from the load  130 , treated now as an input energy source to the bi-directional converter  110 , to generate a current to supply energy to a power supply  120 , treated now as an output load. 
     A bi-directional converter  110  can be configured to receive an input battery voltage at a node  261  from a power supply  120 . In some examples, the power supply  120  can refer to a battery that is located within a device external to the electronic device. A bi-directional converter  110  can be electrically coupled to a load  130  at node  263 . Although  FIG. 2A  shows that the load  130  is modelled as a capacitor C  271 , the load  130  can also refer to an electronic component such as a piezoelectric actuator, an electromechanical actuator, and the like. In another example, the load can be modeled as a capacitor (e.g., capacitor C  271 ) in series with a resistor. 
     The term “reference voltage” can be used interchangeably with the term “reference signal.” In some examples, reference voltage V REF    216  can have a peak value between e.g., 0.5 V to 0.6 V. In some embodiments, reference voltage V REF    216  can have a peak-to-peak range of e.g., 1 V. 
     In conjunction with the bi-directional converter  110  operating in boost mode  281 , the comparator  244  can be configured to determine whether to toggle switch S 1    222  (asserted) with a series of pulses from the PWM unit  291  (actively switching) or leave switch S 1    222  in an open configuration (un-asserted) by comparing V SENSE    217  to the reference voltage V REF    216 . In some examples, the comparator  253  and switching logic  251  can generate a first output value (e.g., CTRL HYS    237 =0, corresponding to an un-asserted switch control) if V SENSE    217  in comparison with V REF    216  does not satisfy a switching threshold value. In some examples, the comparator  253  and switching logic  251  can generate a second output value (e.g., CTRL HYS    237 =1 corresponding to active switching) if V SENSE    217  satisfies the switching threshold value. 
     In conjunction with switch S 1    222  in an active switching configuration that corresponds to, e.g., CTRL HYS =1, the bi-directional converter  110  can be configured to provide energy to sustain a high output load voltage. In some embodiments, the hysteresis control unit  250  introduces a hysteresis amount to avoid excessive repeated pulses of current through the inductor  236 . Excessive repeated pulses of current would follow based on rapid switching of S 1    222 . By applying hysteresis, V OUT    218  follows V REF    216  with some ripple and without excessive switching of S 1    222  and S 2    224 . Examples of CTRL HYS    237  and pulse trains on CTRL 1    235  and CTRL 2    234  and resulting ripple are provided in  FIG. 3 . These examples are provided both for boost mode  281  and buck mode  282 . 
     The hysteresis control unit  250  can be configured to allow a moderate amount of ripple while avoiding excessive switching as the bi-directional converter  110  operates in boost mode  281  or buck mode  282 , independent of a load capacitance, waveform shape, and/or waveform frequency. By implementing an amount of hysteresis that depends on position within a period of V REF    216 , the hysteresis control unit  250  can establish a low voltage transition point and a high voltage transition point that expands beyond the edges of the instantaneous value in time of V REF    216 . In some embodiments, the hysteresis control logic  220  can introduce a positive hysteresis amount (HYSTP) and a negative hysteresis amount (HYSTN). 
     In some embodiments, the PWM unit  291  can be configured to vary the amount of current (I L    203 ) that is provided to the load  130  by switching between a low period and a high period of a pulse duty cycle (see  FIGS. 6A and 6B  for example pulses and currents). In some embodiments, the PWM unit  291  can adjust the duty cycle of the control signals generated in conjunction with boost mode  281  and buck mode  282  to define a minimum and maximum pulse duty cycle so that the shape of the waveform of the load voltage tracks, but does not jump far from, the shape of the waveform of the reference voltage shape. For example, if a fixed wide pulse width to the switch S 1    222  were used, at high values of V REF    216 , the resulting current would allow V OUT    218  to keep up with V REF    216 . However, at low values of V REF    216 , the resulting current would cause V OUT    218  (as represented by V SENSE    217 ) to surge far above V REF    216 . By controlling pulse width as a function of position with a period of V REF    216 , V OUT    218  (as represented by V SENSE    217 ) can be made to track V REF    216  with acceptable ripple. In some examples, the waveform shape of V REF    216 , and thus V OUT    218 , can be sinusoidal or trapezoidal. 
     Output Waveform Illustration, Boost and Buck Modes 
       FIG. 3  illustrates a timing diagram of a modulated waveform of a bi-directional converter  110 , while operating in conjunction with the boost mode  281  and the buck mode  282 , in accordance with some embodiments. The pulse duty cycle and pulse width of the control signals CTRL 1    235  and CTRL 2    234  can be regulated by the PWM unit  291 . In some embodiments, the control logic  220  can be configured to adjust the control signals provided to the PWM unit  291  to adjust the pulse duty cycle. The PWM unit  291  can be configured to regulate an amount of the switching frequency of the switches S 1    222  and S 2    224 . The term “pulse duty cycle” refers to a pulse train, and the fraction of time in the pulse train that a control signal is asserted. The term “pulse width” can refer to a measure of an actual time that an electrical signal associated with the switch is “on” as measured in milliseconds, nanoseconds, and the like. In some examples, the switching frequency of the switches S 1    222  and S 2    224  is higher than the frequency of the output load voltage waveform, V OUT    218 , in order to reduce voltage ripple and generate the cleanest possible spectrum of an output signal. The piece-wise evolution of the V SENSE    217  waveform in  FIG. 3  is provided as a schematic view, and does not represent the underlying current pulses in  FIGS. 6A and 6B  in a scaled fashion. 
     Additionally,  FIG. 3  shows that a shape of the waveform of the output load voltage V OUT    218  substantially corresponds to the shape of the waveform of the reference voltage V REF    216 , which is shown as having a sinusoidal waveform shape. The waveform of the output load voltage follows V REF    216 , and V REF    216  is generated by the waveform generator  292 . 
       FIG. 3  shows a hysteresis range is introduced by the hysteresis control unit  250  so as to introduce positive and negative hysteresis amounts to cause switching to occur at a high voltage transition point (V REF  H) and at a low voltage transition point (V REF  L) instead of at the reference voltage (V REF    216 ). The high voltage transition point (V REF  H) and the low voltage transition point (V REF  L) represent the switching points for the comparator  253 . For example, where the reference voltage is 1 V, the comparator  253  can be configured, via CTRL LEVEL    252 , to establish a high voltage transition point of 1.1 V and a low voltage transition point of 0.9 V. In boost mode  281 , switch S 1    222  initiates active switching or toggling when V SENSE    217  falls below V REF L and continues active switching until V SENSE    217  exceeds V REF  H. Once V SENSE    217  exceeds V REF  H, switch S 1    222  will become un-asserted or open and the energy consumed by the load  130  will cause the V OUT    218  and V SENSE    217  to decease until V SENSE  once again falls below V REF L. In buck mode  282 , switch S 2    224  initiates active switching or toggling when V SENSE    217  is above V REF H and continues active switching (thus generating current I L    203  to the power supply  120 ) until V SENSE    217  falls below V REF  L (due to the energy consumed by load  130 ). As an example, the hysteresis range shown in  FIG. 3  may be 100 mV to 200 mV when the peak to peak voltage of V REF    216  is 1 V. 
       FIG. 3  shows that the boost mode  281  corresponds to V REF    216  and V OUT    218  having a positive slope and the buck mode  282  corresponds to V REF    216  and V OUT    218  having a negative or zero slope. V SENSE    217  in  FIG. 3  is somewhat idealized. Some variation in V SENSE    218  occurs with every pulse on CTRL 1    235  or CTRL 2    234  as explained with respect to  FIGS. 4, 5, 6A, and 6B . 
     In some embodiments, the PWM unit  291  can continually adjust the pulse duty cycle associated with generating the output load voltage and control the amount of voltage ripple at the load  130 . In some examples, an acceptable voltage ripple amount range is e.g., 3%-5% of the total peak voltage. In some embodiments, it may be desirable to have a continually adjusting pulse duty cycle in order to accommodate for a wide range between the low and high voltage transition points. 
     A group of CTRL 1    235  pulses during boost mode  281  is marked as CTRL 1    305 . These occur during a positive comparator value V CMP    254  signal in an instance denoted V H    302 . The pulses CTRL 1    235  are actively switching S 1    222  so that V OUT    218  (represented as V SENSE    217 ) will catch up and exceed V REF    216  to reach V REF H. After V REF    216  exceeds V REF H, V CMP    254  becomes un-asserted (denoted V H    303 ) and the pulse train stops. When V SENSE    217  falls below V REF L, V CMP    254  becomes asserted (an instance denoted as V H    304 ), the pulse train resumes and thus active switching of S 1    222  resumes. 
     A group of CTRL 2    234  pulses during buck mode  282  is marked as CTRL 2    315 . These occur during un-assertion of the comparator V CMP    254  signal in an instance denoted V H    313 . Correspondingly, the switching logic  251  will assert CTRL HYS    237  and the pulses CTRL 2    235  are thus shown actively switching S 2    224  so that V OUT    218  (represented as V SENSE    217 ) will decline below V REF    216  to reach V REF L. During this time, current pulses are flowing into the power supply  120  taking energy from the load  130 . After V REF    216  falls below V REF L, V CMP    254  becomes asserted (denoted V H    314 ) and the pulse train stops. When V SENSE    217  exceeds V REF H, V CMP    254  becomes un-asserted and active switching of S 2    224  resumes. 
     Circuit Diagrams 
       FIGS. 4-5  illustrate circuit diagrams of the bi-directional converter  110  operating in the boost mode  281  and the buck  282  mode, respectively, in accordance with some embodiments. In some embodiments, the hysteresis control unit  250  can be configured to introduce a negative and positive hysteresis amount to establish low and high transition points for operating a bi-directional converter  110  in boost mode  281  and similarly in buck mode  282 . In general, the hysteresis may be asymmetric. That is, V REF  H=V REF    216 +V 1  while V REF L=V REF    216 −V 2 , where V 1  is not equal to V 2 . The hysteresis values V 1  and V 2  may be on the order of 50 mV when V REF    216  has a peak value of 1 V. The hysteresis control unit  250  is configured to continually adjust the hysteresis amount, in accordance with some embodiments. For instance, when a bi-directional converter  110  switches from buck mode  282  to boost mode  281 , the control logic  220  can drive the signal CTRL LEVEL    252  to the hysteresis control unit  250  to adjust a resistor value in the comparator  253 , e.g., and thus change V 1  and/or V 2  from 50 mV to 80 mV. By continually adjusting the hysteresis amount, the hysteresis control unit  250  can reduce voltage ripple. Voltage ripple is exemplified, for example, by the waviness or oscillation of the waveform V SENSE    217  around the waveform V REF    216  visible in  FIG. 3 . 
       FIG. 4  illustrates an exemplary configuration  400  of the bi-directional converter  110  in boost mode  281 . The bi-directional converter  110  is configured to initiate toggling switch S 1    222  to a pulsating or active switching configuration when V SENSE    217 &lt;V REF  L, and stop active switching of switch S 1    222  when V SENSE    217 ≥V REF  H. Switch S 2    224  is configured in an open condition in boost mode  281  and thus CTRL 2    234  is un-asserted. Based on switch S 1    222  closing during a pulse of CTRL 1    235 , I S    212  ramps up as current flows through the inductor  236  to ground (see  FIG. 6A ) and a magnetic field builds up in inductor  236 . When the pulse ends, I S    212  changes quickly to zero, but the current I L    203  continues to flow through the inductor  236 , by the circuit behavior of inductors. I D    221  then begins to flow (solving the summation of currents equation at the node  262 ) and flows into the load  130  represented by C  271  while the magnetic field in the inductor  236  collapses. V OUT    218  then begins to rise, based on integration in a calculus sense, of the current I D    221  in the C  271 . After the magnetic field has collapsed in the inductor  236 , V OUT    218  undergoes little change. The next time a pulse occurs on CTRL 1    235 , the current I S    212  again ramps up while the magnetic field builds in the inductor  236 . When the pulse ends, the magnetic field drives I L    203  and thus the current I D    221  and V OUT    218  is again boosted by accumulation of charge on C  271 . 
       FIG. 5  illustrates an exemplary configuration  500  of the bi-directional converter  110  in buck mode  282 . The bi-directional converter  110  is configured to initiate toggling switch S 2   224  to a pulsating or active switching configuration when VSENSE  217 &gt;VREF H, and stop active switching of switch S 2   224  when VSENSE  217 ≤VREF L. Switch S 1   222  is configured in an open condition in buck mode  282  and thus CTRL 1   235  is un-asserted. Buck mode  282  is not a simple dual of boost mode  281  because the bursts of active switching of S 2   224  are conditioned on what is effectively the input to the circuit charging the power supply  120  at this time. When a pulse arrives on CTRL 2   234 , IS  222  flows through the inductor  236  as IL  203  and into the power supply  120 , this is energy recovery from the load  130 . The sense of direction of IL  203  indicates a negative current at this time, flowing into the power supply  120 . At this time, D 1   215  is back-biased (or reverse biased) and does not turn on. While this flow into power supply  120  is occurring, a magnetic field is building up in the windings of the inductor  236 . When the pulse of CTRL 2   234  ends and S 2   224  opens, a second recharging current flows as ID  211  (solving the current equation at the node  262 ) into the power supply  120 , driven by the collapsing magnetic field in the windings of the inductor  236 . D 2   226  is back-biased and does not turn on. Thus, during the buck mode  282 , two current pulses flow into the power supply  120 , one based on IS  222  and the other based on collapse of the magnetic field in the windings of the inductor  236  when S 2   224  opens. 
     Current Waveforms 
       FIGS. 6A and 6B  give further details of the pulse train instances CTRL 1   305  and CTRL 2   315  introduced in  FIG. 3 .  FIG. 6A  includes y-axis IL  203  and an x-axis of time. Pulses CTRL 1   305  are shown annotated “ON,” referring to the switch S 1   222  being closed. Current flows through the inductor  236  and to ground as IS  212  and is denoted with a bold line on  FIG. 6A . The current waveform is shown somewhat idealized. If the voltage across the inductor  236  is a constant, then the current waveform will be a ramp, that is, a signal proportional to time. The duration of the pulse is shown as w 1   610 . The switch is then opened for a time w 2   611 . The sum of these times is w 3  (reference numeral  612 )=w 1 +w 2 . The duty cycle is w 1 /w 3 . When the switch opens, during the interval of width w 2 , IS  212  must become zero. The magnetic field built up in the windings of the inductor  236  during the time the switch was closed now forces IL  203  through D 2   226  as the current ID  221  (shown as heavy dashed line). This current boosts the voltage on the load  130  modelled as C  271 . The boost occurs because ID  221  causes charge to accumulate on C  271 , and the behavior of a capacitor is such that it integrates, in a calculus sense, current. When the magnetic field has collapsed, the current IL  203  drops to zero until the next pulse arrives, shown with a duration w 4  (reference numeral  614 ). Based on the level of VREF  216  or VOUT  218 , the control logic  220  may cause the PWM unit  291  to create a duration w 4  and period w 5  (reference numeral  615 ) which are different from w 1  and w 3 . 
     With further regard to  FIG. 6A , switch S 1  closes at T 1  and IL  203  ramps up. Switch S 1   222  opens at T 2 . In non-synchronous mode (shown in  FIG. 4 ), IL decays through diode D 2   226  as a function of time. In an alternative embodiment referred to herein as synchronous mode (not shown in  FIG. 6A ), switch S 2   224  closes after switch S 1   222  opens and IL  203  flows through closed switch S 2   224  instead of through diode D 2   226 . Closing switch S 2   224  in combination with the opening of switch S 1   222  may be referred to herein as synchronous mode.  FIG. 6A  does not show this alternative activity on switch S 2   224  carried out by CTRL 2   234 . A small gap in time exists between the opening of switch S 1   222  and the closing of switch S 2   224  so that node  263  is not shorted to ground. Synchronous mode in a switched circuit is also discussed in application Ser. No. 15/421,199 entitle “CHARGER-CONVERTER WITH SINGLE INDUCTOR AND DOWNSTREAM LOW-DROPOUT REGULATOR” filed on Jan. 31, 2017, which is hereby incorporated by reference. Application Ser. No. 15/421,199 and this application are assigned to the same assignee. 
       FIG. 6B  illustrates two currents, I S    222  and I D    211  both of which are buck currents and transfer energy from the load  130  back to the power supply  120 . The sign on I L    203  is (-) to indicate flow towards the power supply  120 . Buck mode  282 , exemplified by the CTRL 2    315  group of pulses, controls the pulse train causing currents to flow toward the power supply  120  acting as an output load. Meanwhile the load  130  is being used as a source or input. However, the control of the bi-directional converter  110  is at the node  262  being used as an input. During the first pulse of CTRL 2    315 , denoted “ON,” switch S 2    224  closes and switch S 1    222  is statically open, as discussed with respect to  FIG. 5 . The current I S    222  builds up (shown as a heavy line) and the magnetic field in the inductor  236  builds up. The current I S    222  is equal to the current IL  203  (based on solving the current equation at the node  262  and neglecting sense of direction), and flows into the power supply  120 . When the pulse ends, the switch S 2    224  opens and the inductor  236  forces current I D    211  through D 1    215 . This current also flows into the power supply  120 , until the magnetic field in the inductor  236  has collapsed. 
     Thus, by means of the bi-directional converter  110 , a circuit with low circuit board area, little heat dissipation, moderate switching activity, and controlled ripple drives a load and periodically recovers energy for a power supply  120  (e.g., a battery) while the output voltage follows a reference voltage and produces the desired effect at the load  130 . 
       FIG. 6B  illustrates a non-synchronous mode in which switch S 1    222  is open and, when switch S 2    224  opens at T 2 , inductor current flows through diode D 1    215 . In order to provide an alternative path for the inductor current I L    203 , a synchronous mode (not shown in  FIG. 6B ) can be used similar to that described with regard to  FIG. 6A . For synchronous mode in  FIG. 6B , after switch S 2    224  opens at T 2 , switch S 1    222  closes until T 3  in order to provide an alternative path for the inductor current I L    203 .  FIG. 6B  does not show this alternative activity on switch S 1    222  carried out by CNRL 1    235 . A small gap in time exists between the opening of switch S 2    224  and the closing of switch S 1    222  so that node  263  is not shorted to ground. 
     Logic 
       FIG. 7  illustrates exemplary logic  700  for operating a bi-directional converter  110  in boost mode  281  and buck mode  282 , in accordance with some embodiments. As shown in  FIG. 7 , the method  700  begins at step  702 , where a comparator  253  receives a fractional load voltage. 
     At step  704 , the comparator  253  (1) determines a slope of the reference voltage, and (2) determines an output value by comparing the fractional load voltage to the reference voltage. 
     At step  706 , the comparator  253  determines whether (1) the slope of the reference voltage is positive, and (2) whether the output value generated by the comparator  253  corresponds to a select output value in conjunction with determining whether to operate the bi-directional converter  232  in boost mode  281 . 
     In some embodiments, the comparator  253  is configured to determine whether the slope (Δvoltage/Δtime) of the reference voltage is either positive (or slope &gt;0) or negative (slope ≤0). The comparator  253  can be configured to generate control signals to the control logic  220  to operate the bi-directional converter  110  in boost mode  281  (step  708 ) when the slope of the reference voltage is positive. 
     Furthermore, in conjunction with the bi-directional converter  110  operating in boost mode  281 , switching logic  251  can be configured to determine whether to actively switch S 1    222  by comparing the fractional load voltage to the reference voltage. Actively switching the switch S 1    222  provides pulses of current to the load  130  to boost the output voltage V OUT    218 . 
     Alternatively, the control logic  220  can be configured to generate control signals to operate the bi-directional converter  110  in buck mode  282  (step  710 ) when the slope of the reference voltage is negative or zero. When the bi-directional converter  110  is operating in buck mode  282 , the control logic  220  can be configured to determine whether to actively switch S 2    224  to cause the output voltage V OUT    218  to track the reference voltage V REF    216  while the power supply  120  is provided with energy from the load  130 . 
     The foregoing description is intended to enable any person skilled in the art to make and use the disclosure, and is provided in the context of a particular application and its requirements. Moreover, the foregoing descriptions of embodiments of the present disclosure have been presented for purposes of illustration and description only. They are not intended to be exhaustive or to limit the present disclosure to the forms disclosed. Accordingly, many modifications and variations will be apparent to practitioners skilled in the art, and the general principles defined herein may be applied to other embodiments and applications without departing from the spirit and scope of the present disclosure. Additionally, the discussion of the preceding embodiments is not intended to limit the present disclosure. Thus, the present disclosure is not intended to be limited to the embodiments shown, but is to be accorded the widest scope consistent with the principles and features disclosed herein.

Metadata:
Filing Date: 20170307
Publication Date: 20190507
Grant Date: 20190507
Priority Date: 20160923
Inventors: REDDICONTO, Salvatore
YANG, DONG
Assignee: APPLE INC
CPC Classifications: [{"code": "H02M3/1582", "inventive": true, "first": true, "tree": "[]"}, {"code": "H02M3/155", "inventive": false, "first": false, "tree": "[]"}, {"code": "H02M1/14", "inventive": true, "first": false, "tree": "[]"}, {"code": "H02M1/14", "inventive": true, "first": false, "tree": "[]"}, {"code": "H02M3/155", "inventive": false, "first": false, "tree": "[]"}, {"code": "H02M3/1582", "inventive": true, "first": true, "tree": "[]"}]
Family ID: 61687242