PATENT DOCUMENT

Publication Number: US-10742118-B2
Application Number: US-201615386340-A
Country: US
Kind Code: B2

Title: Quasi-resonant power converter with reduced dynamic switching losses

Abstract:
A power converter includes a current-monitoring on-state controller that is configured to adjust the timing of a switch-mode voltage conversion stage of the power converter. For example, the timing of the turn-on of a MOSFET associated with a buck converter operated in a discontinuous conduction mode (e.g., quasi-resonant) can be adjusted based on a zero crossing of current through a tank inductor also associated with the buck converter. More particularly, the MOSFET may be turned on after a predetermined delay is initiated after current through the tank inductor reaches zero. The predetermined delay is based on a resonance period defined by the characteristic capacitance of the MOSFET and the inductance of the tank inductor.

Claims:
What is claimed is: 
     
       1. A power converter comprising:
 a step-down voltage converter comprising:
 a tank inductor; 
 an output capacitor in series with the tank inductor; 
 a voltage-controlled switch in series with the tank inductor, the voltage-controlled switch operable in a conducting state and a non-conducting state; and 
 an on-state controller operably connected to the voltage-controlled switch; 
 
 a current monitor comprising:
 an auxiliary coil magnetically coupled with the tank inductor; and 
 a resistor in series with the auxiliary coil; and 
 
 a delay circuit configured to apply a signal to the on-state controller to trigger the conducting state of the voltage-controlled switch after a voltage output from the current monitor crosses a predetermined threshold; wherein
 the signal is applied after a predetermined delay approximately equal to a multiple of a half period of a resonance frequency defined by an inductance of the tank inductor and a capacitance exhibited by the voltage-controlled switch when in the non-conducting state. 
 
 
     
     
       2. The power converter of  claim 1 , wherein the voltage-controlled switch comprises a MOSFET. 
     
     
       3. The power converter of  claim 2 , wherein the delay circuit is configured to initiate the predetermined delay once a voltage across the resistor drops below a threshold. 
     
     
       4. The power converter of  claim 1 , wherein the predetermined delay approximates one period of the resonance frequency. 
     
     
       5. The power converter of  claim 1 , wherein the predetermined delay approximates a future time at which a drain-source voltage of the voltage-controlled switch will drop to a minimum. 
     
     
       6. The power converter of  claim 1 , wherein the current monitor comprises a voltage clamp circuit that sets an upper bound of the voltage across the resistor. 
     
     
       7. The power converter of  claim 1 , wherein the resistor is associated with a voltage divider. 
     
     
       8. The power converter of  claim 1 , wherein:
 the step-down voltage converter is associated with a first ground loop; and 
 the output capacitor defines an output voltage node associated with a second ground loop. 
 
     
     
       9. The power converter of  claim 8 , wherein the first ground loop is isolated from the second ground loop by a galvanic isolation. 
     
     
       10. The power converter of  claim 9 , wherein the galvanic isolation comprises an opto-coupler. 
     
     
       11. The power converter of  claim 8 , wherein the output voltage node is coupled to a snubbing filter. 
     
     
       12. The power converter of  claim 1 , wherein the power converter is configured to be enclosed within a housing of an electronic device. 
     
     
       13. The power converter of  claim 1 , wherein the step-down voltage converter is coupled to an output of a full-bridge rectifier that receives alternating current from an alternating current voltage source. 
     
     
       14. The power converter of  claim 1 , wherein the predetermined delay is approximately one and a half periods of the resonance frequency. 
     
     
       15. A power converter within a housing of an electronic device, the power converter comprising:
 a resonant buck converter comprising a tank inductor connected in series with a MOSFET; 
 a current monitoring circuit magnetically coupled with the tank inductor; and 
 an on-state controller configured to increase a gate voltage of the MOSFET above a switching threshold of the MOSFET, after a predetermined delay approximately equal to a multiple of a half period of a resonance frequency defined by an inductance of the tank inductor and a capacitance of the on-state controller, when a voltage output from the current monitoring circuit crosses a predetermined threshold. 
 
     
     
       16. The power converter of  claim 15 , wherein:
 the predetermined delay corresponds to the resonance frequency associated with the tank inductor and the capacitance; and 
 the capacitance is exhibited by the MOSFET when the MOSFET is in a non-conducting state. 
 
     
     
       17. The power converter of  claim 15 , wherein the current monitoring circuit comprises:
 an auxiliary inductor magnetically coupled with the tank inductor; 
 a voltage divider in series with the auxiliary inductor; and 
 a voltage clamp coupled to the voltage divider and configured to set an upper bound of a voltage output from the voltage divider. 
 
     
     
       18. A method of reducing voltage in a power converter, the method comprising:
 receiving a voltage at an input of a buck converter comprising a tank inductor and a voltage-controlled switch; 
 monitoring current through an auxiliary inductor isolated from the buck converter; 
 upon determining that the current through the auxiliary inductor has dropped to zero, initiating a predetermined delay; and 
 after the predetermined delay, increasing a switching voltage of the voltage-controlled switch; wherein 
 the predetermined delay is approximately equal to a multiple of a half of one period of a resonance frequency defined by the tank inductor and a capacitance exhibited across the voltage-controlled switch when current through the voltage-controlled switch is zero. 
 
     
     
       19. The method of  claim 18 , wherein the predetermined delay is based, at least in part, on a switching frequency of the buck converter. 
     
     
       20. The method of  claim 19 , wherein the predetermined delay is selected such that the voltage-controlled switch is turned on when instantaneous power consumption of the voltage-controlled switch is at a minimum.

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This application is a non-provisional patent application of, and claims the benefit to, U.S. Provisional Patent Application No. 62/370,603, filed Aug. 3, 2016, and titled “Quasi-Resonant Power Converter With Reduced Dynamic Switching Losses,” the disclosure of which is hereby incorporated herein by reference in its entirety. 
    
    
     FIELD 
     Embodiments described herein generally relate to power converters and, in particular, to systems and methods for reducing switching losses in power converters. 
     BACKGROUND 
     An electronic device can receive electric power from a power source. The electronic device can include a power conversion and/or regulation circuit to change one or more characteristics of power received from the power source into a form usable by one or more components of the electronic device. In many examples, the power conversion and/or regulation circuit includes a power converter, such as a buck converter, boost converter, or a boost-buck converter. 
     A reduction in the physical size of a power converter incorporated within, or associated with, an electronic device is often desired. In these cases, power converters are conventionally implemented with smaller output capacitors and are conventionally operated at higher switching frequencies. However, as the operational frequency of a power converter is increased, dynamic switching losses accumulate more rapidly and operational efficiency of the power converter decreases. 
     SUMMARY 
     Embodiments described herein generally reference a power converter including a step-down voltage converter. The step-down voltage converter can include a tank inductor, an output capacitor in series with the tank inductor, a voltage-controlled switch in series with the tank inductor, and an on-state controller operably connected to the voltage-controlled switch. The on-state controller is coupled to a current monitor including an auxiliary coil magnetically coupled with the tank inductor. In this manner, when a voltage output from the current monitor crosses a predetermined threshold, a signal can be applied to the on-state controller after a predetermined delay (referred to herein as a “delayed signal”). The delayed signal turns the voltage-controlled switch on at a time that reduces switching losses. 
     Other embodiments described herein generally reference a power converter disposed within a housing of an electronic device. The power converter includes a resonant buck converter formed with a tank inductor connected in series with a MOSFET. In addition, the power converter includes a current monitoring circuit magnetically coupled with the tank inductor. The power converter also includes an on-state controller configured to increase a gate voltage of the MOSFET, after a predetermined delay, when a voltage output from the current monitoring circuit crosses a predetermined threshold. In other words, the current monitoring circuit is configured to detect the condition when current through the tank inductor reaches zero amps. 
     Still further embodiments described herein generally reference a method of reducing voltage in a power converter, the method including the operations of: receiving a voltage at an input of a buck converter defined by a tank inductor and a voltage-controlled switch and monitoring current through an auxiliary inductor isolated from the buck converter. Upon determining that current through the auxiliary inductor has dropped to zero amps, a predetermined delay is initiated, after which a switching voltage of the voltage-controlled switch is raised. In this manner, turn-on of the voltage-controlled switch is delayed until voltage across leads of the voltage-controlled switch when in the off-state is minimized (e.g., zero voltage switching). In these embodiments, the predetermined delay can be based on a resonance frequency associated with the tank inductor and a capacitance exhibited across the voltage-controlled switch when the voltage-controlled switch is not conducting. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Reference will now be made to representative embodiments illustrated in the accompanying figures. It should be understood that the following descriptions are not intended to limit this disclosure to one preferred embodiment. To the contrary, the disclosure provided herein is intended to cover alternatives, modifications, and equivalents as may be included within the spirit and scope of the described embodiments, and as defined by the appended claims. 
         FIG. 1A  depicts an electronic device coupled to a stand-alone power converter. 
         FIG. 1B  depicts a stand-alone power converter configured for use with a wireless power transfer system. 
         FIG. 1C  depicts another stand-alone power converter configured for use with a wireless power transfer system. 
         FIG. 2  is a simplified system diagram of a power converter that incorporates a step-down voltage converter such as described herein. 
         FIG. 3  is a simplified system diagram of a power converter that incorporates a step-down voltage converter and a current-monitoring on-state controller, such as described herein. 
         FIG. 4A  is a simplified schematic diagram of a rectifier of a power converter, such as described herein. 
         FIG. 4B  is a simplified schematic diagram of a buck converter of a power converter, such as described herein. 
         FIG. 4C  is a simplified schematic diagram of a feedback controller of a power converter, such as described herein. 
         FIG. 4D  is a simplified schematic diagram of a low-pass filter network of a power converter, such as described herein. 
         FIG. 4E  is a simplified schematic diagram of a current-monitoring on-state controller of a power converter, such as described herein. 
         FIG. 5  is a simplified schematic diagram of a power converter incorporating a current-monitoring on-state controller, such as described herein. 
         FIG. 6A  is a simplified signal diagram tracking a voltage level that may be used to toggle an on-state of a buck converter of a power converter, such as described herein. 
         FIG. 6B  is a simplified signal diagram tracking current through a tank inductor of the buck converter described in reference to  FIG. 6A . 
         FIG. 6C  is a simplified signal diagram tracking voltage across a switch of the buck converter described in reference to  FIG. 6A . 
         FIG. 7  is a simplified flow chart depicting example operations of a method of operating a current-monitoring on-state controller, such as described herein. 
     
    
    
     The use of the same or similar reference numerals in different figures indicates similar, related, or identical items. 
     The use of cross-hatching or shading in the accompanying figures is generally provided to clarify the boundaries between adjacent elements and also to facilitate legibility of the figures. Accordingly, neither the presence nor the absence of cross-hatching or shading conveys or indicates any preference or requirement for particular materials, material properties, element proportions, element dimensions, commonalities of similarly illustrated elements, or any other characteristic, attribute, or property for any element illustrated in the accompanying figures. 
     Additionally, it should be understood that the proportions and dimensions (either relative or absolute) of the various features and elements (and collections and groupings thereof) and the boundaries, separations, and positional relationships presented therebetween, are provided in the accompanying figures merely to facilitate an understanding of the various embodiments described herein and, accordingly, may not necessarily be presented or illustrated to scale, and are not intended to indicate any preference or requirement for an illustrated embodiment to the exclusion of embodiments described with reference thereto. 
     DETAILED DESCRIPTION 
     Embodiments described herein reference systems and methods for operating a power converter in a manner that efficiently converts one or more characteristics of electric power received from an electric power source (more generally, “power source”) into a form usable by one or more components of an electronic device. The electronic device may be any stationary or mobile electronic device such as a desktop computer, a laptop computer, a tablet computer, a cellular telephone, a peripheral device, an accessory device, a wearable device, a vehicle or aeronautical control system, an industrial control system, an appliance, and so on. 
     Generally, a power converter, such as described herein, is configured to convert voltage from an unregulated or otherwise noisy voltage source (herein, “input voltage”) into a regulated voltage level (herein, “output voltage”) suitable for use by one or more electronic devices. For example, a power converter can be configured to regulate mains voltage (e.g., 90 VAC-265 VAC at 50-60 Hz) to a reference level such as 3.3 VDC, 5.0 VDC, 12 VDC, 50 VDC or any other suitable reference voltage. In some examples, the output of the power converter can be boosted to a higher level after being regulated to the reference level. 
     For simplicity and consistency of the description provided herein, many embodiments are presented and described with reference to power converters configured to reduce a high voltage alternating current (e.g., 265 VAC) to a relatively lower voltage direct current (e.g., 50 VDC). It may be appreciated, however, that the various techniques, circuit topologies, operations and/or methods presented with respect to this particular implementation can be equivalently applied to power converters configured to regulate power in another manner. For example, a power converter such as described herein can be suitably configured in any implementation-specific manner to convert an arbitrary input voltage to any selected or desired output voltage, whether such operation requires DC-to-DC conversion stages, AC-to-DC conversion stages, DC-to-AC conversion stages, AC-to-AC conversion stages, or any combination or sequence thereof. 
     As noted above, some embodiments described herein reference a power converter configured to reduce an input voltage level to a particular output voltage level. In these examples, the power converter includes at least one step-down voltage converter, such as a buck converter, operated at a duty cycle selected to efficiently reduce the input voltage to the output voltage level. In many cases, the output of the step-down voltage converter is connected to a feedback circuit, such as a compensation network. The output of the step-down voltage converter can thereafter be connected to a load, such as an electronic device. 
     In some cases, a power converter may be constrained to a small size. As a result of reduced size, the step-down voltage converter may be required to operate at a higher frequency. However, high frequency operation of a step-down voltage converter conventionally results in increased dynamic switching losses and reduced power efficiency. 
     Accordingly, some embodiments described herein reference a power converter that incorporates a current-monitoring on-state controller (a “current monitor”) configured to influence the turn-on timing of a step-down voltage converter that is configured to operate at high frequency. More specifically, the current-monitoring on-state controller includes an auxiliary inductor that is configured to measure current through one or more portions of the step-down voltage converter. Once current through the auxiliary inductor, and thus the monitored portion of the step-down voltage converter, crosses a threshold (e.g., 0.0 A for a step-down voltage converter operating in discontinuous-conduction mode), the current-monitoring on-state controller can trigger a turn-on of the step-down voltage converter. 
     In further embodiments, the current-monitoring on-state controller is coupled in series with a delayed trigger. In these embodiments, once current through the auxiliary inductor crosses a threshold, the current-monitoring on-state controller causes the delayed trigger to initiate a delay. The delay is typically based on a characteristic resonance frequency of one or more components of the step-down voltage converter. The delay can be a predetermined delay or can be dynamically changed. In this manner, the time at which the step-down voltage converter is turned on can be delayed until voltage across the voltage-controlled switch approaches or reaches a minimum value, such as zero volts. In this manner, dynamic switching losses (of the voltage-controlled switch) are minimized or eliminated and operational power efficiency of the power converter is increased. 
     These and other embodiments are discussed below with reference to  FIGS. 1A-7 . However, those skilled in the art will readily appreciate that the detailed description given herein with respect to these figures is for explanation only and should not be construed as limiting. 
     Generally and broadly,  FIGS. 1A-1C  reference various example electronic devices that may incorporate, or may be associated with or coupled to, one or more power converters such as described herein. It will be appreciated, however, that the depicted examples are not exhaustive; the various embodiments described with reference to  FIGS. 1A-1C  may be modified or combined in any number of suitable or implementation-specific ways. 
     For example,  FIG. 1A  depicts an electronic device coupled to a stand-alone power converter configured to change one or more characteristics of power received from a power source into a form usable by the electronic device. 
     More particularly, the electronic device  100   a  includes a housing  102  to retain, support, and/or enclose various components of the electronic device  100   a  such as a rechargeable battery (not shown). The electronic device  100   a  can also include a processor, memory, power converter and/or battery, network connections, sensors, input/output ports, acoustic elements, haptic elements, digital and/or analog circuits for performing and/or coordinating tasks of the electronic device  100   a , and so on. For simplicity of illustration, the electronic device  100   a  is depicted in  FIG. 1A  without many of these elements, each of which may be included, partially and/or entirely, within the housing  102  and may be operationally or functionally associated with the internal battery. 
     In one example, the internal battery of the electronic device  100   a  can be recharged by physically connecting the electronic device  100   a  to a power converter  104 . More specifically, a power cable  106  can provide a direct electrical connection between the power converter  104  and the electronic device  100   a . In some cases, the power cable  106  is connected to the electronic device  100   a  via a connector  108 . 
     In these embodiments, the power converter  104  can be configured to accept power at mains voltage and output that power in a form usable by one or more circuits configured to facilitate recharging of the internal battery. In one particular example, the power converter  104  accepts 120 VAC as input and outputs 5 VDC, which can be accepted by the electronic device  100   a  and used to recharge the internal battery. More broadly, the power converter  104  can be configured to accept high-voltage AC and can be configured to output a lower-voltage DC. 
     In another example, the power converter  104  can be configured to accept power at mains voltage and output that power in a form that is subsequently converted again by the electronic device  100   a  prior to being used to charge the internal battery. More specifically, in this example, the power converter  104  can be configured to accept 120 VAC as input and can be configured to output 50 VDC. In these examples, the power converter  104  may also include an inverter (not shown). Thereafter, the electronic device  100   a  can accept 50 VDC and further convert, by a second power converter within the electronic device  100   a , to 5 VDC. 
     More broadly, the power converter  104  can be configured in this example to accept high-voltage AC and can be configured to output lower-voltage DC. In addition, the second power converter (which can be enclosed within the housing  102 ) can be configured to accept relatively high-voltage DC and can be configured to output low-voltage DC. 
     It may be appreciated that the limited examples provided above are not exhaustive. For example, the power converter  104  may be configured to perform an AC-to-AC or AC-to-DC conversion to different voltages than those provided above. Similarly, a power converter enclosed within the housing of the electronic device  100   a  may be appropriately configured to provide AC-to-AC, AC-to-DC, DC-to-AC, or DC-to-DC conversion. 
     Furthermore, although illustrated as a cellular phone, it may be appreciated that the electronic device  100   a  can be another suitable electronic device that is either stationary or mobile, taking a larger or smaller form factor than illustrated. For example, in certain embodiments, the electronic device  100   a  can be a laptop computer, a tablet computer, a cellular phone, a wearable device, a health monitoring device, a home or building automation device, a home or building appliance, a craft or vehicle entertainment, control, power, and/or information system, a navigation device, and so on. 
     In still further embodiments, a power converter (such as the power converter  104 ) can be configured to operate with an inductive or resonant wireless power transfer system. For example,  FIG. 1B  depicts a stand-alone power converter, identified as the power converter  100   b , configured to change one or more characteristics of power received from a power source into a form that may be wirelessly transferred to an electronic device (not shown). In this example, the power converter  100   b  can be configured to convert alternating current received via a connector end  110  into alternating current (at the same or different frequency) that can be used by a transmitter end  112  of the power converter  100   b  to generate one or more time-varying magnetic fields that can be used to wirelessly transfer power to an electronic device placed on or near the transmitter end  112 . In this example, the power converter  100   b  can directly convert alternating current at one frequency and peak-to-peak voltage into altering current at a second frequency and peak-to-peak voltage. In this manner, the power converter  100   b  may operate more efficiently; an intermediate conversion to direct current is not required. 
     Another example of an inductive or resonant wireless power transfer system is depicted in  FIG. 1C . More specifically, a power converter  100   c  can be configured to change one or more characteristics of power received from a power source into a form that may be wirelessly transferred to more than one electronic device is illustrated. As with the example described above, the power converter  100   c  can be configured to convert alternating current received via a connector end  110  into alternating current (at the same or different frequency) that can be used by a transmitter end  114  of the power converter  100   c  to generate one or more time-varying magnetic fields that can be used to wirelessly transfer power to multiple electronic devices each placed on or near the transmitter end  114 . In this example, the power converter  100   c  directly converts alternating current at one frequency and peak-to-peak voltage into altering current at a second frequency and peak to peak voltage. In this manner, and as noted with respect to some embodiments described herein, the power converter  100   c  may operate more efficiently; an intermediate conversion to direct current is not required. 
     The foregoing embodiments depicted in  FIGS. 1A-1C  and the various alternatives thereof and variations thereto are presented, generally, for purposes of explanation, and to facilitate an understanding of various possible electronic devices or accessory devices that can incorporate, or be otherwise coupled to, one or more power converters such as described herein. More specifically,  FIGS. 1A-1C  are presented to illustrate that a power converter such as described herein can be incorporated, either entirely or partially, into the housing of an electronic device, into a separate power accessory that couples to an electronic device via a cable, into a separate power accessory that provides wireless power to one or more electronic devices, and so on. 
     Generally and broadly,  FIGS. 2-3  reference a power converter that may be incorporated within, coupled to or otherwise associated with, an electronic device such as the electronic devices depicted in  FIGS. 1A-1C . 
     For example,  FIG. 2  depicts a simplified system diagram of a power converter that incorporates a step-down voltage converter such as described herein. The power converter  200  can be configured to accept power from an alternating current power source  202 . 
     The alternating current power source  202  can deliver alternating current with any suitable amplitude or frequency. In one example, the alternating current power source  202  is connected to the output of a step-up converter (not shown) which can be configured to accept variable mains voltage as input (e.g., 110 VAC-265 VAC). In this case, the step-up converter may be configured to increase mains voltage to 400 VDC, or any other suitable voltage level that is reliably higher than the maximum expected mains voltage level (e.g., 265 VAC). 
     In other examples, the alternating current power source  202  can be implemented in another way. For example, the alternating current power source  202  can be a receive coil (or more than one receive coil) of a wireless power transfer system. More specifically, an inductive or resonant power transfer system can include a transmit coil and a receive coil that, when positioned in proximity of one another, form a primary coil and secondary coil of an air-gap transformer. When power transfers from the primary coil to the secondary coil, the secondary coil outputs alternating current. 
     However, for simplicity of illustration and description, the embodiments that follow are described in reference to an alternating current power source  202  configured to output high voltage alternating current, such as 90 VAC or 265 VAC, although as noted with respect to some embodiments described herein (e.g., wireless power transfer embodiments), any suitable alternating current power source can be stepped down or otherwise adjusted using the techniques, methods, and circuit topologies described below. 
     The power converter  200  can include multiple distinct and interconnected circuits or blocks, such as, but not limited to: a rectifier  204 , a step-down voltage converter  206 , and a compensation network  208 . In some cases, the power converter  200  can also include or be associated with one or more of a processor, memory, sensors, digital and/or analog circuits for performing and/or coordinating tasks of the power converter  200 . For simplicity of illustration and description, the power converter  200  is depicted in  FIG. 2  without many of these elements. 
     The rectifier  204  of the power converter  200  receives alternating current from the alternating current power source  202  and rectifies the received current into a rippled direct current. The rectifier  204  can be a half-bridge rectifier, although in many embodiments a full-bridge rectifier is used. In many cases, a filter can be added in parallel to the output of the rectifier to further smooth the rippled direct current waveform. The filter can be any suitable low-pass filter (e.g., a capacitor or capacitor network parallel to the output of the rectifier or an inductor-capacitor choke or filter). The rectifier  204  can be implemented in any number of suitable ways. For example, the rectifier  204  can be a synchronous or asynchronous rectifier. For embodiments in which the alternating current power source  202  outputs ˜265 VAC, the rectifier  204  outputs rippled direct current having an average bias of (up to) 400 VDC. 
     The step-down voltage converter  206  of the power converter  200  receives the rippled direct current from the rectifier  204 . In many embodiments, the step-down voltage converter  206  is implemented with a buck converter topology. 
     In one example, a buck converter can include a tank inductor and an output capacitor. A low-side lead of the tank inductor is coupled to a high-side lead of the output capacitor, which, in turn, is connected in parallel to an output ground lead of the buck converter. The output leads of the buck converter are typically connected to a load. In many cases, a compensation network or other regulator is positioned between the output leads of the buck converter and the input leads of the load. The compensation network can provide regulation and ripple smoothing to the voltage received by the load. For simplicity of illustration, these components are not shown in  FIG. 2 . 
     A return diode couples a low-side lead of the output capacitor of the buck converter to a high-side lead of the tank inductor. The buck converter also includes a voltage-controlled switch that couples the high-side lead of the tank inductor to an input lead of the buck converter. The input lead of the buck converter receives the input voltage, which in the illustrated example is the rippled direct current output from the rectifier  204 . 
     The buck converter can be switched between an on-state and an off-state by toggling the voltage-controlled switch. The buck converter topology described above is referred herein as a “high-side” buck converter as a consequence of the direct connection between the voltage-controlled switch and the input voltage received from the rectifier  204 . 
     When a high-side buck converter is in the on-state, the voltage-controlled switch is closed and a first current loop is defined from the input voltage source, through the tank inductor, to the load. In this state, voltage across the tank inductor sharply increases to a voltage level equal to the difference between the voltage across the load and the input voltage. This voltage across the tank inductor induces current through the tank inductor to linearly increase. As a result of the topology of the circuit, the current flowing through the tank inductor also flows to the output capacitor and to the load. 
     Alternatively, when the high-side buck converter transitions to the off-state, the voltage-controlled switch is opened and a second current loop is defined through the return diode. In this state, voltage across the tank inductor sharply decreases to a voltage level equal to the difference between the voltage across the output leads of the buck converter and the cut-in voltage of the return diode. This voltage across the tank inductor is lower than when in the on-state, so current within the tank inductor linearly decreases in magnitude. The decreasing current flowing through the tank inductor also flows to the output capacitor and to the load connected across the output leads of the buck converter. In this manner, the output capacitor functions as a low-pass filter, generally reducing ripple in the voltage delivered from the output of the buck converter to the load. 
     The buck converter can be efficiently operated by switching between the on-state and the off-state by toggling the voltage-controlled switch at a duty cycle selected based on the desired output voltage. The voltage output from the buck converter is proportionately related to the input voltage by the duty cycle. This relationship can be modeled by Equation 1: 
     
       
         
           
             
               
                 
                   
                     D 
                     cycle 
                   
                   = 
                   
                     
                       V 
                       out 
                     
                     
                       V 
                       in 
                     
                   
                 
               
               
                 
                   Equation 
                   ⁢ 
                   
                       
                   
                   ⁢ 
                   1 
                 
               
             
           
         
       
     
     In one example, if direct current output from the rectifier  204  is 400 VDC and the desired output voltage is 50 VDC, a duty cycle of 12.5% may be selected. 
     In many cases, the buck converter is operated in a discontinuous conduction mode, although this may not be required. More particularly, if the buck converter is operated in a discontinuous conduction mode, current through the tank inductor regularly reaches 0.0 A. In some embodiments, the buck converter can be operated at or near resonance frequency of the tank inductor and the output capacitor. 
     In still further embodiments, the step-down voltage converter  206  can be implemented in another manner; it is appreciated that the example topology described above is merely one example of a suitable or appropriate step-down voltage converter. 
     For example, in another embodiment, the high-side lead of the tank inductor is coupled to a low-side lead of the output capacitor, which, in turn, is connected in parallel to the load. The return diode couples a low-side lead of the tank inductor to a high-side lead of the output capacitor. The voltage-controlled switch couples the low-side lead of the tank inductor to a ground reference of the buck converter. This topology is referred to herein as a “low-side” buck converter as a consequence of the connection between the voltage-controlled switch and the input voltage ground reference. In some cases, a step-down voltage converter  206  may be implemented with a low-side buck converter in order to have the same ground reference between the rippled direct current ground (connected to the load) and the output ground of the step-down voltage converter  206 . 
     In many examples, the output of the step-down voltage converter  206  of the power converter  200  is rippled direct current having a voltage defined by the duty cycle at which the step-down voltage converter  206  is operated. In many cases, rippled direct current may not be preferred. As such, the compensation network  208  (or other voltage regulator) may be positioned between the output of the step-down voltage converter  206  and the load. The compensation network  208  can be configured to reduce ripple within the voltage output from the step-down voltage converter  206  and, additionally, provide feedback to a shunt voltage regulator coupled to the step-down voltage converter  206  so that the duty cycle of the step-down voltage converter  206  can be adjusted in real-time. 
     As noted with respect to some embodiments described herein, it may be required to operate the step-down voltage converter  206  at a high frequency. In this case, the power converter  200  incorporates a current-monitoring on-state controller (not shown) that is configured to adjust the turn-on timing of the step-down voltage converter  206 . More specifically, the current-monitoring on-state controller typically includes an auxiliary inductor (see, e.g.,  FIG. 3 ) that is configured to measure current through the tank inductor of the step-down voltage converter  206 . More particularly, the auxiliary inductor is positioned so as to be magnetically coupled to the tank inductor. Once current through the auxiliary inductor, and thus the tank inductor, reaches zero amps a resonance may be caused between the tank inductor and a stray capacitance across a voltage-controlled switch within the step-down voltage converter  206 . The current-monitoring on-state controller can trigger a turn-on of the voltage-controlled switch when voltage across the voltage-controlled switch is at a minimum (e.g., at or near zero volts). In this manner, the current-monitoring on-state controller facilitates quasi-resonant operation of the step-down voltage converter  206 . 
     In many cases, the current-monitoring on-state controller incorporates or is coupled to a delayed trigger (see, e.g.,  FIG. 3 ). In these embodiments, once current through the auxiliary inductor crosses a threshold, the current-monitoring on-state controller causes the delayed trigger to initiate a delay. The delay is typically based on a characteristic resonance frequency of the inductor and the characteristic capacitance of the voltage-controlled switch when the switch is open. In this manner, the time at which the step-down voltage converter  206  is turned on can be delayed until voltages associated with those components in resonance are at a minimum value (see, e.g.,  FIGS. 3, 6A-6C ). In this manner, dynamic switching losses are minimized and/or eliminated, and operational power efficiency of the power converter  200  is increased. 
       FIG. 3  depicts a simplified system diagram of another example power converter, identified as the power converter  300 . The power converter  300  can include multiple distinct and interconnected circuits or blocks, such as, but not limited to: a voltage level converter  304 , a compensation network or regulator (identified as the regulator  306 ), and a current-monitoring on-state controller  320 . As with some embodiments described herein, the power converter  300  can also include or be associated with one or more of a processor, memory, sensors, and digital and/or analog circuits for performing and/or coordinating tasks of the power converter  300 . For simplicity of illustration and description, the power converter  300  is depicted in  FIG. 3  without many of these elements. 
     The voltage level converter  304  of the power converter  300  receives the rippled direct current from a rippled direct current source  302 . In many embodiments, the voltage level converter  304  is implemented with a low-side buck converter topology. For example, a low-side buck converter includes a tank inductor  310  and an output capacitor  312 . In this topology, the high-side lead of the tank inductor  310  is coupled to a low-side lead of the output capacitor  312 , which, in turn, is connected to the load  314  via the regulator  306 . 
     A return diode (not shown) couples a low-side lead of the tank inductor  310  to a high-side lead of the output capacitor  312 . The return diode defines a second current path through the tank inductor  310  to the load  314  when the buck converter is in an off-state. A MOSFET  316  can be used to couple the low-side lead of the tank inductor  310  to a ground reference of the buck converter. 
     The voltage level converter  304 , implemented as a low-side buck converter, can be switched between an on-state and an off-state by toggling the MOSFET  316 . More specifically, when the low-side buck converter is in the on-state, the MOSFET  316  is closed and a first current loop is defined from the rippled direct current source  302 , through the tank inductor  310 , to the load  314 . In this state, voltage across the tank inductor  310  sharply increases to a voltage level equal to the difference between the voltage across the load  314  and the input voltage. This voltage across the tank inductor  310  induces current through the tank inductor  310  to linearly increase. As a result of the topology of the circuit, the current flowing through the tank inductor  310  also flows to the output capacitor  312  and to the load  314 . 
     Alternatively, when the low-side buck converter transitions to the off-state, the MOSFET  316  is opened and a second current loop is defined through the return diode. In this state, voltage across the tank inductor  310  sharply decreases to a voltage level equal to the difference between the voltage across the load  314  and the cut-in voltage of the return diode. This voltage across the tank inductor  310  is lower than when in the on-state, so current within the tank inductor  310  linearly decreases in magnitude. The decreasing current flowing through the tank inductor  310  also flows to the output capacitor  312  and to the load  314  connected across the output leads of the buck converter. In this manner, the output capacitor  312  functions as a low-pass filter, generally reducing ripple in the voltage delivered from the output of the buck converter to the load  314 . The regulator  306  further reduces remaining ripple in the voltage signal. 
     As noted with respect to some embodiments described herein, the buck converter can be efficiently operated by switching between the on-state and the off-state by toggling the MOSFET  316  at a duty cycle selected based on the desired output voltage. The MOSFET  316  can be controlled by an on-state trigger  318 . The on-state trigger  318  is configured to selectively apply a voltage to the gate of the MOSFET  316  to enable the on-state of the buck converter. 
     As noted with respect to some embodiments described herein, it may be required to operate the voltage level converter  304  at a high frequency. In this case, the power converter  300  incorporates a current-monitoring on-state controller  320  that is configured to adjust the turn-on timing of the voltage level converter  304  by controlling a time at which the on-state trigger  318  is enabled. More specifically, the current-monitoring on-state controller  320  typically includes an auxiliary inductor  322  that is positioned adjacent to the tank inductor  310 . In this manner, current through the auxiliary inductor  322  is equal to current through the tank inductor  310 . As noted above, once current through the auxiliary inductor  322 , and thus the tank inductor  310  reaches zero amps, a resonance between the tank inductor  310  and a capacitance exhibited by the MOSFET  316  (in a non-conducting state) can be caused. To mitigate switching losses during this resonance, the current-monitoring on-state controller  320  can trigger a turn-on of the voltage level converter  304  by waiting (e.g., triggering a predetermined delay after which the gate voltage of the MOSFET  316  may be raised) until the drain-source voltage of the MOSFET  316  becomes zero or near-zero. In this manner, the current-monitoring on-state controller  320  facilitates quasi-resonant operation of the voltage level converter  304 . 
     More specifically, the current-monitoring on-state controller  320  can include a voltage divider  324  that is appropriately coupled in parallel or series with the auxiliary inductor  322 . In this manner, as current is induced through the auxiliary inductor  322  by the tank inductor  310 , voltage across the voltage divider  324  can change. More specifically, voltage across a resistor of the voltage divider  324  can be changed. As a result of the clamping, the voltage as measured from a measurement node of the voltage divider  324  can increase as current through the auxiliary inductor  322  decreases. This voltage signal can be used to control the on-state timing of the MOSFET  316 . In this manner, generally and broadly, the current-monitoring on-state controller  320  can be used to facilitate zero-current and/or zero-voltage switching of the MOSFET  316  which, in turn, can reduce dynamic switching losses. 
     In further cases, the current-monitoring on-state controller  320  can also (optionally) incorporate a delay block  326  (e.g., a circuit configured to provide a signal, after a predetermined delay, in response to an input signal). The delay block  326  can be inserted between the output of the voltage divider  324  and the on-state trigger  318 . The duration provided by the delay block  326  can be typically based on a characteristic resonance frequency of the tank inductor  310  and the characteristic capacitance of the MOSFET  316 , when open. In this manner, the time at which the voltage level converter  304  is turned on can be delayed until voltages associated with those components in resonance are at a minimum value (see, e.g.,  FIGS. 6A-6C ). In this manner, dynamic switching losses are further minimized and/or eliminated, and operational power efficiency of the power converter  300  is increased. 
     The foregoing embodiments depicted in  FIGS. 2-3  and the various alternatives thereof and variations thereto are presented, generally, for purposes of explanation, and to facilitate a thorough understanding of various possible configurations and circuit topologies of a power converter that incorporates a current-monitoring on-state controller that facilitates efficient quasi-resonant operation. However, it will be apparent to one skilled in the art that some of the specific details presented herein may not be required in order to practice a particular described embodiment, or an equivalent thereof. 
     Generally and broadly,  FIGS. 4A-5  reference certain distinct and interconnected purpose-configured circuits that can be used to implement a power converter such as described herein. It will be appreciated, however, that the depicted examples are not exhaustive; the various embodiments depicted and described with reference to  FIGS. 4A-5  may be implemented, interconnected, or otherwise modified in any number of suitable or appropriate ways. 
     For example,  FIG. 4A  depicts a simplified schematic diagram of an example rectifier of a power converter such as described herein. The rectifier  400  receives alternating current from an alternating current power source, (such as the alternating current power source  202  shown in  FIG. 2 ) and rectifies the received current into a rippled direct current. More particularly, the rectifier  400  can be coupled to a two leads of an alternating current power source, identified as nodes V AC_pos  and V AC_neg  which are respectively associated with the rectifier input lead  402   a  and the rectifier input lead  402   b.    
     The rectifier  400  can be a half-bridge rectifier, although in the illustrated embodiment a full-bridge rectifier formed by the diode network  404  is used. In many cases, a filter capacitor  406  can be added in parallel to the output V rect  of the rectifier  400  to further smooth the rippled direct current waveform. The filter capacitor  406  functions as a low-pass filter. Although the rectifier is illustrated as a passive full-wave rectifier, such a configuration may not be required of all embodiments. For example, the rectifier  400  can be a synchronous rectifier in which the diodes of the diode network  404  are substituted for switches such as MOSFETs. The output terminal of the rectifier  400 , which is configured to output rippled direct current (e.g., rectified alternating current), is identified as the rectifier output terminal  408 . 
       FIG. 4B  depicts a simplified schematic diagram of a high-side buck converter of a power converter such as described herein. As with other buck converters described herein, the high-side buck converter  410  includes an output capacitor  412  and a tank inductor  414 . In this topology, the high-side lead of the tank inductor  414  is coupled to a low-side lead of the output capacitor  412 , which, in turn, is connected to output leads of the high-side buck converter  410 . Specifically, output leads of the high-side buck converter  410  include a positive output terminal  416  and a negative output terminal  418 . 
     A return diode  420  couples a low-side lead of the tank inductor  414  to a high-side lead of the output capacitor  412 . The return diode  420  defines a second current path through the tank inductor  414  to the positive output terminal  416  when the high-side buck converter  410  is in an off-state. A MOSFET  422  can be used to couple the low-side lead of the tank inductor  414  to a ground reference of the high-side buck converter  410 . 
     The voltage level converter, implemented as a high-side buck converter  410 , can be switched between an on-state and an off-state by toggling the MOSFET  422 . More specifically, when the high-side buck converter  410  is in the on-state, the MOSFET  422  is closed and a first current loop is defined from the rectifier output terminal  408 , through the tank inductor  414 , to the positive output terminal  416 . 
     In the on-state, voltage across the tank inductor  414  sharply increases to a voltage level equal to the difference between the voltage across the output capacitor  412  and the input voltage V rect . This voltage across the tank inductor  414  induces current i L  through the tank inductor  414  to linearly increase. As a result of the topology of the circuit, the current i L  flowing through the tank inductor  414  also flows to the output capacitor  412  and to the positive output terminal  416 . 
     When the high-side buck converter  410  transitions to the off-state, the MOSFET  422  is opened and a second current loop is defined through the return diode  420 . In this state, voltage across the tank inductor  414  sharply decreases to a voltage level equal to the difference between the voltage across the output capacitor  412  and the cut-in voltage of the return diode  420  (e.g., 0.7 VDC). This voltage across the tank inductor  414  is lower than when in the on-state, so the current i L  within the tank inductor  414  linearly decreases in magnitude. The decreasing current i L  flowing through the tank inductor  414  also flows to the output capacitor  412  and to the positive output terminal  416  connected across the output leads of the high-side buck converter  410 . In this manner, the output capacitor  412  functions as a low-pass filter, generally reducing ripple in the voltage delivered from the output of the high-side buck converter  410  to the positive output terminal  416 . 
     More particularly, as noted with respect to some embodiments described herein, the high-side buck converter  410  can be efficiently operated by switching between the on-state and the off-state by toggling the gate voltage V gate  of the MOSFET  422  at a duty cycle selected based on the desired output voltage. More specifically, increasing the voltage at the gate  424  of the MOSFET  422  can cause the MOSFET  422  to conduct current. In many embodiments, a voltage at the source  426  of the MOSFET  422 , V source  can be used as input to a comparator discussed in reference to  FIG. 4C , discussed below. 
     The high-side buck converter  410  can be toggled from the on-state to the off-state in a manner that is responsive to changes in impedance of a load. More particularly, the high-side buck converter  410  can include a feedback controller  428 , such as shown in  FIG. 4C . The feedback controller  428  can include a flip-flop  430  that is reset via output from a comparator  432 . 
     The comparator  432  receives negative input by probing an output of a compensation network at an isolated node  434  to determine a feedback voltage V feedback  (see, e.g.,  FIG. 4D ). The comparator  432  receives positive input by probing the voltage V source  at the source  426  of the MOSFET  422  (see  FIG. 4B ) when the MOSFET  422 . The comparator  432  compares the difference between the negative input V feedback  and the positive input V source  to produce an output. 
     When the MOSFET  422  is in the on-state and is conducting current, the resistor  422   a  is connected to the source of the MOSFET  422 . As a result, voltage across the resistor  422   a  corresponds to current through the MOSFET  422 . As the MOSFET  422  begins conducting current, voltage across the resistor  422   a  increases. This increase in voltage across the resistor  422   a  is fed to the positive input of the comparator  432  (see, e.g.,  FIG. 4C ). The negative input of the comparator  432  is received from the voltage control loop, V feedback . As noted above, the output of the comparator  432  is connected to the reset input of the flip-flop  430 . As a result of the depicted configuration, the flip-flop  430  will be reset (e.g., the output of the comparator  432  will be high) when V sense  is larger than V feedback . When the flip-flop  430  is reset, the MOSFET  422  will be turned off. Thereafter, the voltage of the source of the MOSFET  422  will be zero volts and current through the resistor  422   a  will be zero amps (e.g., because the MOSFET  422  is not conducting current from drain to source). The set input of the flip-flop  430  is fed from a fixed clock frequency; after the MOSFET  422  turns off, the next few switching cycles will be triggered by the fixed clock frequency. 
     In this manner, if the output voltage of the power converter drops, V feedback  increases which, in turn, will increase the per-cycle on-time of the MOSFET  422  because the flip-flop  430  will retain the set condition until the voltage across the resistor  422   a  equals the higher feedback voltage. As may be appreciated, a higher voltage across the resistor  422   a  corresponds to a higher current through the MOSFET  422  (e.g., higher peak current) which increases the output voltage of the power converter. In this manner, the output voltage of the power converter can be regulated independent of the load current and independent of the input voltage. The flip-flop  430  also includes a set lead  436  which is discussed in further detail in reference to  FIG. 4E . 
     The output of the high-side buck converter  410  can be coupled to a compensation network and/or a regulation network in order to regulate V OUT_BUCK  against any load variation.  FIG. 4D  depicts a simplified schematic diagram of one such network that includes, among other components, an output voltage feedback network with loop compensation to stabilize a dynamic response of the system in case of load variation. 
     The compensating network  438  can be formed in any number of suitable ways to provide stable operation of the voltage output from the high-side buck converter  410  and to provide an isolated node suitable for providing feedback to the comparator  432 . 
     More particularly, the compensating network  438  as shown in the illustrated embodiment includes two input terminals, two output terminals, and the isolated node  434  for feedback to the comparator  432 . The two input terminals are a negative output terminal  416  and a positive output terminal  418  of the high-side buck converter  410  (see, e.g.,  FIG. 4B ). The two output terminals are identified as the positive output load terminal  440  and the output ground reference terminal  456 . 
     The compensating network  438  can include an opto-coupler  444  that can be used to isolate (e.g., to support galvanic isolation) the isolated node  434  and to isolate a ground loop associated with the high-side buck converter  410  from a ground loop associated with the output of the power converter. A high side of the light-emitting portion of the opto-coupler  444  can be coupled to the input terminal of the compensating network  438 , which has a potential equal to V OUT_BUCK . In this manner, changes in the value of V OUT_BUCK  (e.g., ripple), can result in changes in brightness of the opto-coupler  444 , which in turn can change the voltage at the isolated node  434  used by the comparator  432  to control the duration of the on-state of the MOSFET  422 . 
     It may be required to operate the high-side buck converter  410  at a high frequency. In this case, the power converter can incorporate a current-monitoring on-state controller that is configured to adjust the turn-on timing of the high-side buck converter  410  by controlling a time at which the on-state trigger is enabled. In some cases, a bias resistor  446  can be positioned parallel to the light-emitting portion of the opto-coupler  444  to provide bias current to the shunt regulator  450 . 
     The compensating network  438  can also include a resistor-capacitor network  448  which can be suitably tuned and/or designed in order to provide stable feedback to the comparator  432  via the opto-coupler  444 . Particularly, the resistor-capacitor network  448  can be coupled to a shunt voltage regulator  450  that provides a constant voltage reference based on a voltage divider defined by the of a resistor  452   a  and the resistor  452   b.    
     The compensating network  438  can also include a resistor-capacitor network  457  to sense output voltage and/or current. In the illustrated example, the resistor-capacitor network  457  can operate to stabilize the V OUT_BUCK  signal. The resistor-capacitor network  457  is typically coupled in parallel with the resistor  452   a.    
     As noted with respect to some embodiments described herein, it may be required to operate the high-side buck converter  410  at a high frequency. In this case, as shown in  FIG. 4E , the power converter incorporates a current-monitoring on-state controller  458  that is configured to adjust the turn-on timing of the high-side buck converter  410  by controlling a time at which the flip-flop  430  is set. More specifically, the current-monitoring on-state controller  458  typically includes an auxiliary inductor  460  that is positioned adjacent to the tank inductor  414 . 
     In this manner, the auxiliary inductor  460  is configured to directly detect zero or near-zero current through the tank inductor  414 . In addition, the auxiliary inductor  460  is configured to indirectly detect the condition at which drain-source voltage of the MOSFET  422  is zero or near-zero. As noted above, once current through the auxiliary inductor  460 , and thus the tank inductor  414 , reaches zero amps, a resonance is caused between the MOSFET  422  and the tank inductor  414 , which can lead to switching losses. Therefore, in these embodiments, the current-monitoring on-state controller  458  can trigger a turn-on of the MOSFET  422  by raising the voltage of the set lead  436  of the flip-flop  430  only after the resonance condition causes voltage across the MOSFET  422  to be zero or near-zero (e.g., one period of resonance, a half period of resonance, or any other suitable time). In this manner, the current-monitoring on-state controller  458  facilitates quasi-resonant operation of the high-side buck converter  410 . 
     More specifically, the current-monitoring on-state controller  458  can include a voltage divider  462  that is appropriately coupled in parallel or series with the auxiliary inductor  460 . In this manner, as current is induced through the auxiliary inductor  460  by the tank inductor  414 , the auxiliary inductor  460  can be used to detect zero current through the tank inductor  414  by monitoring the voltage across the voltage divider  462 . 
     As a result, the voltage as measured from a measurement node of the voltage divider  462  can be used to detect zero or near-zero current through the MOSFET  422  and zero or near-zero drain-source voltage across the MOSFET  422 . As noted with respect to some embodiments described herein, this voltage signal can be used to control the on-state timing of the MOSFET  422 . In this manner, generally and broadly, the current-monitoring on-state controller  458  can be used to facilitate zero-current and/or zero-voltage switching of the MOSFET  422  which, in turn, can reduce dynamic switching losses. 
     In further cases, the current-monitoring on-state controller  458  can also incorporate a delay block  466 . In some embodiments, the delay block  466  is an inverter delay circuit. In other cases, the delay block  466  can be configured to provide a variable delay. One of ordinary skill in the art would recognize that various analog and digital circuits, or combinations thereof, can provide delayed output signal(s) responsive to input signal(s). 
     The delay block  466  is inserted between the output of the voltage divider  462  and the set lead  436  of the flip-flop  430 . The duration provided by the delay block  466  can be typically based on a characteristic resonance frequency of the tank inductor  414  and the characteristic capacitance of the MOSFET  422 , when open. 
     More specifically, when the MOSFET  422  is in the on-state and current is flowing through the tank inductor  414 , voltage across the MOSFET  422  is effectively 0.0V (e.g., disregarding conduction losses). However, when the MOSFET  422  is in the off-state, a voltage between the drain and the source  426  exists; the MOSFET  422  exhibits a capacitance. In the off-state, as noted above, current i L  through the tank inductor  414  drops linearly toward zero amps and the return diode  420  stops conducting. As a result, capacitance within the MOSFET  422  may resonate with the tank inductor  414 , which can induce a current of opposite sign to flow through the tank inductor  414 . If the MOSFET  422  were to be turned on when the voltage across the source  426  and the drain were greater than zero, switching losses would occur. Thus, the delay provided by the delay block  466  is selected so that the MOSFET  422  is turned on only after the voltage across the drain and source of the MOSFET  422  reaches a valley or minimum in the resonance period. 
     In this manner, the time at which the high-side buck converter  410  is turned on can be delayed until voltages associated with those components in resonance are at a minimum value (see, e.g.,  FIGS. 6A-6C ). In this manner, dynamic switching losses are further minimized and/or eliminated, and operational power efficiency of the power converter is increased. 
       FIG. 5  depicts a simplified schematic diagram of a power converter that may be implemented with the various schematics depicted and described in reference to  FIGS. 4A-4E . In particular, the power converter  500  is configured to receive high-voltage AC and configured to output reduced voltage DC. Particularly, the power converter  500  includes a rectifier  502  that feeds into a high-side buck converter  504 . The output of the high-side buck converter  504  can be regulated by a compensating/feedback network  506 . Feedback received from the compensating/feedback network  506  can be supplied to a comparator  508 . Finally, the power converter  500  can include a current-monitoring on-state controller  510  that assists in timing the turn-on of a MOSFET of the high-side buck converter  504  so that parasitic capacitances that develop and resonate within the power converter  500  during the off-state of the buck converter do not exacerbate switching losses. 
     The foregoing embodiments depicted in  FIGS. 4A-5  and the various alternatives thereof and variations thereto are presented, generally, for purposes of explanation, and to facilitate a thorough understanding of various possible configurations of circuits that may be used to implement a power converter such as described herein. However, it will be apparent to one skilled in the art that some of the specific details presented herein may not be required in order to practice a particular described embodiment, or an equivalent thereof. 
     Generally and broadly,  FIGS. 6A-6C  are simplified signal diagrams corresponding to steady state operation of a power converter such as described herein. 
     For example,  FIG. 6A  depicts a simplified signal diagram tracking a voltage level that may be used to toggle an on-state of a buck converter (see, e.g.,  FIG. 4B ) of a power converter such as described herein. In particular, the signal diagram  600  depicts a gate voltage that may be used to control the conduction of a voltage-controlled switch (e.g., MOSFET, IGBT, and so on), such as the MOSFET  422  shown in  FIG. 4B . In the illustrated example, the gate voltage is toggled at a duty cycle approximately equal to 33%. More specifically, the gate voltage is on from time t 0  to time t 1  and off from time t 1  to time t 2 ; the associated voltage-controlled switch is in a conducting state from time t 0  to time t 1 . 
       FIG. 6B  depicts a simplified signal diagram  602  tracking current through a tank inductor (e.g., the tank inductor  414  as shown in  FIG. 4B ) of the buck converter described in reference to  FIG. 6A . As noted with respect to some embodiments described herein, current through the tank inductor of a buck converter increases linearly if a constant voltage is applied across the voltage 
               (       e   .   g   .     ,       dI   dt     =     V   L         )     .         
Thus, the current through the tank inductor increases as the time approaches t 1 . At time t 1 , the gate voltage of the voltage-controlled switch shifts to 0.0V; the voltage controlled switch stops conducting. At this time, voltage across the tank inductor reduces to a value equivalent to the output voltage of the buck converter less the cut-in voltage of a return diode. As described above, this change in voltage causes current through the tank inductor to linearly decrease. If the buck converter is operated in a discontinuous conduction mode, the current can approach and cross 0.0 A, identified in  FIG. 6B  as the zero crossing  604 .
 
     As noted with respect to some embodiments described herein, a power converter can include a current-monitoring on-state controller that includes an auxiliary inductor positioned nearby the tank inductor of a buck converter such that the auxiliary inductor and the tank inductor are magnetically coupled. As such, it may be appreciated that the current depicted in  FIG. 6B  can also equivalently represent the current through an auxiliary inductor of a current-monitoring on-state controller. 
       FIG. 6C  depicts a simplified signal diagram  606  tracking voltage across the voltage-controlled switch of the buck converter described in reference to  FIG. 6A . As noted with respect to some embodiments described herein, a potential difference exists across the voltage-controlled switch (e.g., the drain-source voltage of a MOSFET, such as the MOSFET  422  shown in  FIG. 4B ) when the voltage-controlled switch is not conducting. In the illustrated embodiment, the voltage-controlled switch stops conducting at time t.sub. 1 , after which a voltage is present across the voltage-controlled switch. Once current through the inductor (tank inductor or auxiliary inductor) drops to zero volts at the zero crossing  604 , the voltage across the voltage-controlled switch can resonate with the tank inductor, as shown in  FIG. 6C . The period of this resonance is based on the characteristic capacitance of the voltage-controlled switch and the inductance of the tank inductor. In this example, the voltage-controlled switch may be turned on again after a delay  607  that is based on the resonance period. In this manner, the voltage-controlled switch is turned on when the voltage across the voltage-controlled switch is at a minimum. This can prevent further resonance and power loss and/or switching losses, which are represented as the avoided losses  608 . In this manner, dynamic switching losses are minimized and/or eliminated, and operational power efficiency of the power converter is increased. Additionally, one of skill in the art will appreciate that the delay  607  need not be based on a single period of resonance; in other embodiments the voltage-controlled switch can be turned off at a first valley of resonance (e.g., half a period of resonance), at a second valley of resonance (e.g., a full resonance period), at a third valley of resonance, and so on. 
       FIG. 7  is a simplified flow chart depicting example operations of a method of operating a current-monitoring on-state controller such as described herein. The method depicted can, in some embodiments, be performed (at least in part) by one or more portions of a power converter such as depicted in  FIGS. 4A-5 . In other cases, the method is performed by another processor or circuit, or combination of processors or circuits. 
     The method  700  begins at operation  702  in which a zero crossing of current is detected. In some cases, an auxiliary inductor can be positioned adjacent to a tank inductor of a buck converter. Current through the auxiliary inductor is equal to current through the tank inductor. In other cases, a different current can be measured. For example, current can be measured by an inductor or Hall effect sensor positioned around or adjacent a conductor, such as a lead or trace associated with a buck converter of a power converter. Accordingly, it may be appreciated that a zero crossing of current can be measured or obtained in any number of suitable ways. 
     Next, at operation  704 , a delay is initiated. The delay can be based, at least in part, on a resonance frequency associated with the power converter. In one specific example, the resonance frequency is based on a characteristic capacitance of a MOSFET and the inductance of a tank inductor that are each associated with a buck converter. More specifically, the delay can be configured to approximate one period of the resonance frequency, thereby approximating a future time at which the drain-source voltage of the MOSFET is at a minimum. Lastly, at operation  706 , the MOSFET can be enabled. 
     One may appreciate that although many embodiments are disclosed above, that the operations and steps presented with respect to methods and techniques described herein are meant as exemplary and accordingly are not exhaustive. One may further appreciate that alternate step order or fewer or additional operations may be required or desired for particular embodiments. 
     Although the disclosure above is described in terms of various exemplary embodiments and implementations, it should be understood that the various features, aspects and functionality described in one or more of the individual embodiments are not limited in their applicability to the particular embodiment with which they are described, but instead can be applied, alone or in various combinations, to one or more of the some embodiments of the invention, whether or not such embodiments are described and whether or not such features are presented as being a part of a described embodiment. Thus, the breadth and scope of the present invention should not be limited by any of the above-described exemplary embodiments but is instead defined by the claims herein presented.

Metadata:
Filing Date: 20161221
Publication Date: 20200811
Grant Date: 20200811
Priority Date: 20160803
Inventors: OH, InHwan
Assignee: APPLE INC
CPC Classifications: [{"code": "H02M1/0058", "inventive": false, "first": false, "tree": "[]"}, {"code": "H02M1/0058", "inventive": false, "first": false, "tree": "[]"}, {"code": "H02M1/14", "inventive": true, "first": false, "tree": "[]"}, {"code": "Y02B70/10", "inventive": false, "first": false, "tree": "[]"}, {"code": "H02M1/14", "inventive": true, "first": false, "tree": "[]"}, {"code": "H02M3/156", "inventive": true, "first": true, "tree": "[]"}, {"code": "H02M3/156", "inventive": true, "first": true, "tree": "[]"}, {"code": "Y02B70/10", "inventive": false, "first": false, "tree": "[]"}, {"code": "H02M3/156", "inventive": true, "first": true, "tree": "[]"}, {"code": "Y02B70/1425", "inventive": false, "first": false, "tree": "[]"}, {"code": "Y02B70/1491", "inventive": false, "first": false, "tree": "[]"}, {"code": "H02M2001/0058", "inventive": false, "first": false, "tree": "[]"}, {"code": "H02M1/14", "inventive": true, "first": false, "tree": "[]"}]
Family ID: 61069632