PATENT DOCUMENT

Publication Number: US-8031094-B2
Application Number: US-55837409-A
Country: US
Kind Code: B2

Title: Touch controller with improved analog front end

Abstract:
A controller for a touch sensor includes a transimpedance amplifier, and a feedback resistor coupled to an input of the transimpedance amplifier and to an output of the transimpedance amplifier. At least one multiplexor may be coupled to the input of the transimpedance amplifier and configured to multiplex a plurality of analog inputs to one dedicated channel. The controller may further include a bandpass filter coupled to the output of the transimpedance amplifier. The output of the bandpass filter may be input to an anti-aliasing filter, which feeds into an analog to digital converter. Alternatively, the output of the bandpass filter may be input to a sigma-delta analog to digital converter.

Claims:
1. An analog front end for receiving a sense signal from a touch sensor, comprising:
 a transimpedance amplifier; 
 a feedback resistor coupled to an input of the transimpedance amplifier and to an output of the transimpedance amplifier; and 
 a bandpass filter coupled to the output of the transimpedance amplifier; 
 wherein a gain of the bandpass filter is adjustable to increase noise headroom. 
 
     
     
       2. The analog front end of  claim 1 , wherein the bandpass filter has a programmable center frequency. 
     
     
       3. The analog front end of  claim 1 , further comprising:
 an analog to digital converter coupled to the output of the bandpass filter. 
 
     
     
       4. The analog front end of  claim 3 , wherein the analog to digital converter is a sigma-delta analog to digital converter. 
     
     
       5. The analog front end of  claim 3 , further comprising:
 an anti-aliasing filter coupled between the output of the bandpass filter and the analog to digital converter. 
 
     
     
       6. The analog front end of  claim 5 , wherein a cut-off frequency of the anti-aliasing filter can be adjusted based on a selected center frequency of the bandpass filter as to optimize bandpass filter roll-off. 
     
     
       7. The analog front end of  claim 1 , further comprising:
 at least one multiplexor coupled to the input of the transimpedance amplifier configured to multiplex a plurality of sense signals to one analog front end. 
 
     
     
       8. The analog front end of  claim 1 , wherein the touch sensor is part of a computing system. 
     
     
       9. The analog front end of  claim 1 , wherein the bandpass filter has a passband gain of at least 6 dB. 
     
     
       10. A method for receiving a sense signal from a touch sensor, comprising:
 amplifying an incoming signal using a transimpedance amplifier having a feedback resistor coupled to an input of the transimpedance amplifier and to an output of the transimpedance amplifier; and 
 filtering an output of the transimpedance amplifier using a bandpass filter having a programmable center frequency; and 
 adjusting a gain of the bandpass filter to increase noise headroom. 
 
     
     
       11. The method of  claim 10 , further comprising:
 digitizing an output of the bandpass filter to enable digital demodulation. 
 
     
     
       12. The method of  claim 11 , wherein the analog to digital converter is a sigma-delta analog to digital converter. 
     
     
       13. The method of  claim 10 , further comprising:
 attenuating noise by performing anti-aliasing filtering of an output of the bandpass filter. 
 
     
     
       14. The method of  claim 13 , wherein a cut-off frequency of the anti-aliasing filtering can be adjusted based on the programmable center frequency so as to optimize bandpass filter roll-off. 
     
     
       15. The method of  claim 10 , further comprising:
 multiplexing a plurality of analog inputs to one dedicated channel to be input to the transimpedance amplifier. 
 
     
     
       16. The method of  claim 10 , wherein the touch sensor controller is part of a computing system. 
     
     
       17. The method of  claim 10 , wherein the bandpass filter has a passband gain of at least 6 dB. 
     
     
       18. A controller for a touch sensor, comprising:
 a pre-amplifier; 
 a feedback resistor coupled to an input of the pre-amplifier, an output of the pre-amplifier and a virtual ground; 
 at least one capacitor coupled to the input of the pre-amplifier and the virtual ground; 
 a sigma-delta analog to digital converter coupled to the output of the pre-amplifier; and 
 a feedback circuit coupled between an output of the sigma-delta analog to digital converter and the input of the pre-amplifier to provide variable gain adjustment and bandpass filter response. 
 
     
     
       19. The controller of  claim 18 , wherein the sigma-delta analog to digital converter is of a second order or higher. 
     
     
       20. The controller of  claim 18 , wherein two capacitors are coupled to the input of the pre-amplifier, with at least one capacitor coupled to virtual ground and at least one capacitor has a variable capacitance. 
     
     
       21. A controller for a touch sensor, comprising:
 a pre-amplifier, the input of which is coupled to a virtual ground; and 
 a sigma-delta analog to digital converter coupled to the output of the pre-amplifier, wherein the sigma-delta analog to digital converter has bandpass filter response, with one or a plurality of feedback paths to the input of the pre-amplifier, which is held at virtual ground. 
 
     
     
       22. The controller of  claim 21 , wherein the bandpass filter response is realized with a resonator. 
     
     
       23. The controller of  claim 21 , wherein the sigma-delta analog to digital converter is of a second order or higher. 
     
     
       24. The controller of  claim 21 , further comprising:
 a digital demodulator coupled to the output of the sigma-delta analog to digital converter. 
 
     
     
       25. An analog front end for receiving a sense signal from a touch sensor, comprising:
 a biquad bandpass filter including a virtual ground input for rejecting stray capacitance from the touch sensor; and 
 an analog to digital converter (ADC) coupled to the biquad bandpass filter. 
 
     
     
       26. The analog front end of  claim 25 , where the biquad bandpass filter has at least one of a programmable center frequency, gain and Q. 
     
     
       27. The analog front end of  claim 25 , where the ADC is a sigma-delta convertor. 
     
     
       28. The analog front end of  claim 25 , where the biquad bandpass filter is a tow-thomas biquad bandpass filter.

Description:
FIELD 
     This relates generally to touch sensor panels, and in particular, to touch controllers with improved analog front ends. 
     BACKGROUND 
     Many types of input devices are presently available for performing operations in a computing system, such as buttons or keys, mice, trackballs, joysticks, touch sensor panels, touch screens and the like. Touch screens, in particular, are becoming increasingly popular because of their ease and versatility of operation as well as their declining price. Touch screens can include a touch sensor panel, which can be a clear panel with a touch-sensitive surface, and a display device such as a liquid crystal display (LCD) that can be positioned partially or fully behind the panel so that the touch-sensitive surface can cover at least a portion of the viewable area of the display device. Touch screens can allow a user to perform various functions by touching the touch sensor panel using a finger, stylus or other object at a location dictated by a user interface (UI) being displayed by the display device. In general, touch screens can recognize a touch event and the position of the touch event on the touch sensor panel, and the computing system can then interpret the touch event in accordance with the display appearing at the time of the touch event, and thereafter can perform one or more actions based on the touch event. 
     Mutual capacitance touch sensor panels can be formed from a matrix of drive and sense lines of a substantially transparent conductive material such as Indium Tim Oxide (ITO), often arranged in rows and columns in horizontal and vertical directions on a substantially transparent substrate. Drive signals can be transmitted through the drive lines, which can result in the formation of static mutual capacitance at the crossover points (sensing pixels) of the drive lines and the sense lines. The static mutual capacitance, and any changes to the static mutual capacitance due to a touch event, can be determined from sense signals that can be generated in the sense lines due to the drive signals. 
     SUMMARY 
     This relates to a touch controller for a touch sensor panel that can have improved charge handling capability, noise immunity, and a smaller footprint. The touch controller can include a transimpedance amplifier in its analog front end for receiving a sense signal from a touch sensor panel, with a feedback resistor coupled to an input of the transimpedance amplifier and to an output of the transimpedance amplifier. The transimpedance amplifier can provide improved noise attenuation in the analog front end, while consuming less DIE real estate, as compared to a charge amplifier with capacitive feedback. The gain of the transimpedance amplifier can be programmable via a programmable feedback resister. At least one multiplexor may be coupled to the input of the transimpedance amplifier and configured to multiplex a plurality of analog inputs to one dedicated channel, in order to consume even less DIE real-estate. 
     The analog front end can further include a bandpass filter coupled to the output of the transimpedance amplifier. The bandpass filter can increase available noise headroom in the output of the transimpedance amplifier. The center frequency of the bandpass filter can be programmable. The output of the bandpass filter can be input to an anti-aliasing filter, which feeds into an analog to digital converter. The cutoff frequency of the anti-aliasing filter can be programmable. Alternatively, the output of the bandpass filter can be input directly to a sigma-delta analog to digital converter, in which case the anti-aliasing filter may be unnecessary. 
     An alternate embodiment is directed to an analog front end for a touch sensor, which can include a pre-amplifier, with a feedback resistor coupled to an input of the pre-amplifier, an output of the pre-amplifier and a virtual ground. The controller can further include at least one capacitor coupled to the input of the pre-amplifier and the virtual ground, and a sigma-delta analog to digital converter coupled to the output of the pre-amplifier. 
     Yet another embodiment is directed to an analog front end for a touch sensor, which can include a pre-amplifier, the input of which can be coupled to a virtual ground; and a sigma-delta analog to digital converter can be coupled to the output of the pre-amplifier. The sigma-delta analog to digital converter can have band-pass filter response and can have one or more feedback paths from the sigma-delta converter to the pre-amplifiers input which can be held at virtual ground. The center frequency of the sigma-delta delta converter&#39;s integrated bandpass filter can be programmable. 
     Yet another embodiment is directed to an analog front end for a touch sensor which can include a biquad bandpass filter, whose input can be held at virtual ground; and a sigma delta analog to digital converter can be coupled to the output of the biquad bandpass filter. The biquad bandpass filter can have a programmable center frequency. 
     Yet another embodiment is directed to an analog front end for a touch sensor which can include a biquad bandpass filter, whose input can be held at virtual ground; and an anti-aliasing filter can be coupled to the output of the biquad band-pass filter and the output of the anti-aliasing filter can be coupled the output of an analog to digital converter. The analog to digital converter can be a successive approximation or pipeline analog to digital converter. The anti-aliasing filter can have a programmable cut-off frequency. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The present disclosure, in accordance with one or more various embodiments, is described in detail with reference to the following figures. The drawings are provided for purposes of illustration only and merely depict exemplary embodiments of the disclosure. These drawings are provided to facilitate the reader&#39;s understanding of the disclosure and should not be considered limiting of the breadth, scope, or applicability of the disclosure. It should be noted that for clarity and ease of illustration these drawings are not necessarily made to scale. 
         FIG. 1  illustrates an example computing system according to various embodiments. 
         FIG. 2   a  illustrates an exemplary mutual capacitance touch sensor panel according to various embodiments. 
         FIG. 2   b  is a side view of an exemplary pixel in a steady-state (no-touch) condition according to various embodiments. 
         FIG. 2   c  is a side view of an exemplary pixel in a dynamic (touch) condition according to various embodiments. 
         FIG. 3  illustrates an example application-specific integrated circuit (ASIC) single chip multi-touch controller according to various embodiments. 
         FIG. 4  illustrates details of one of the sense channels and digital demodulation section according to various embodiments. 
         FIG. 5  illustrates an exemplary single-ended analog front end, with a transimpedance amplifier (TIA), a bandpass filter (BPF), a anti-aliasing filter (AAF) and an analog-to-digital converter (ADC), according to various embodiments. 
         FIG. 6  illustrates an exemplary single-ended analog front end with a transimpedance amplifier (TIA), a bandpass filter (BPF) and a Sigma-Delta ADC, according to various embodiments. 
         FIG. 7  illustrates an exemplary transimpedance amplifier (TIA), according to various embodiments. 
         FIG. 8  illustrates an exemplary bandpass filter in an analog front end, according to various embodiments. 
         FIG. 9  illustrates an exemplary anti-aliasing filter in an analog front end, according to various embodiments. 
         FIG. 10  illustrates exemplary frequency response plots, according to various embodiments. 
         FIG. 11  illustrates an exemplary analog front end combining transimpedance amplifier (TIA), bandpass and analog to digital converter (ADC) functions into a single block, according to various embodiments. 
         FIG. 12  illustrates a high level concept of a biquad filter, according to various embodiments. 
         FIG. 13  illustrates an exemplary Tow Thomas biquad filter, according to various embodiments. 
         FIG. 14   a  illustrates an exemplary mobile telephone that can include a touch sensor panel according to the various embodiments described herein. 
         FIG. 14   b  illustrates an exemplary digital media player that can include a touch sensor panel according to the various embodiments described herein. 
         FIG. 14   c  illustrates exemplary personal computer that can include a touch sensor panel according to the various embodiments described herein. 
     
    
    
     DETAILED DESCRIPTION 
     In the following description of embodiments, reference is made to the accompanying drawings which form a part hereof, and in which it is shown by way of illustration specific embodiments that can be practiced. It is to be understood that other embodiments can be used and structural changes can be made without departing from the scope of the disclosed embodiments. 
     This relates to a touch controller for a touch sensor panel that can have improved charge handling capability, noise immunity, and a smaller footprint. The touch controller can include a transimpedance amplifier in its analog front end for receiving a sense signal from a touch sensor panel, with a programmable feedback resistor coupled to an input of the transimpedance amplifier and to an output of the transimpedance amplifier. At least one multiplexor can be coupled to the input of the transimpedance amplifier and configured to multiplex a plurality of analog inputs to one dedicated channel. The controller can further include a bandpass filter coupled to the output of the transimpedance amplifier. The output of the bandpass filter can be input to an anti-aliasing filter, which feeds into an analog to digital converter. Alternatively, the output of the bandpass filter can be input to a sigma-delta analog to digital converter. 
     It should be understood that the various embodiments can be applicable to both mutual and self-capacitance sensor panels, single and multi-touch sensor panels, and other sensors in which multiple simultaneous stimulation signals are used to generate a composite sense signal. Furthermore, it should be understood that various embodiments can also be applicable to various touch sensor panel configurations, such as configurations in which the drive and sense lines are formed in non-orthogonal arrangements, on the back of a cover glass, on the same side of a single substrate, or integrated with display circuitry. 
       FIG. 1  illustrates example computing system  100  that can utilize touch controller  106  with integrated drive system according to various embodiments. Touch controller  106  can be a single application specific integrated circuit (ASIC) that can include one or more processor subsystems  102 , which can include, for example, one or more main processors, such as ARM968 processors or other processors with similar functionality and capabilities. However, in other embodiments, the processor functionality can be implemented instead by dedicated logic, such as a state machine. Processor subsystems  102  can also include, for example, peripherals (not shown) such as random access memory (RAM) or other types of memory or storage, watchdog timers and the like. Touch controller  106  can also include, for example, receive section  107  for receiving signals, such as touch sense signals  103  from the sense lines of touch sensor panel  124 , other signals from other sensors such as sensor  111 , etc. Touch controller  106  can also include, for example, a demodulation section such as multistage vector demod engine  109 , panel scan logic  110 , and a drive system including, for example, transmit section  114 . Panel scan logic  110  can access RAM  112 , autonomously read data from the sense channels and provide control for the sense channels. In addition, panel scan logic  110  can control transmit section  114  to generate stimulation signals  116  at various frequencies and phases that can be selectively applied to the drive lines of touch sensor panel  124 . 
     Charge pump  115  can be used to generate the supply voltage for the transmit section. Stimulation signals  116  (Vstim) can have amplitudes higher than the maximum voltage the ASIC process can tolerate by cascoding transistors. Therefore, using charge pump  115 , the stimulus voltage can be higher (e.g. 6V) than the voltage level a single transistor can handle (e.g. 3.6 V). Although  FIG. 1  shows charge pump  115  separate from transmit section  114 , the charge pump can be part of the transmit section. 
     Touch sensor panel  124  can include a capacitive sensing medium having a plurality of drive lines and a plurality of sense lines. The drive and sense lines can be formed from a transparent conductive medium such as Indium Tin Oxide (ITO) or Antimony Tin Oxide (ATO), although other transparent and non-transparent materials such as copper can also be used. In some embodiments, the drive and sense lines can be perpendicular to each other, although in other embodiments other non-Cartesian orientations are possible. For example, in a polar coordinate system, the sensing lines can be concentric circles and the driving lines can be radially extending lines (or vice versa). It should be understood, therefore, that the terms “drive lines” and “sense lines” as used herein are intended to encompass not only orthogonal grids, but the intersecting traces of other geometric configurations having first and second dimensions (e.g. the concentric and radial lines of a polar-coordinate arrangement). The drive and sense lines can be formed on, for example, a single side of a substantially transparent substrate, opposite sides of the same substrate, or on two different substrates. 
     At the “intersections” of the traces, where the drive and sense lines can pass adjacent to and above and/or below (cross) each other (but without making direct electrical contact with each other), the drive and sense lines can essentially form two electrodes (although more than two traces could intersect as well). Each intersection of drive and sense lines can represent a capacitive sensing node and can be viewed as picture element (pixel)  126 , which can be particularly useful when touch sensor panel  124  is viewed as capturing an “image” of touch. (In other words, after touch controller  106  has determined whether a touch event has been detected at each touch sensor in the touch sensor panel, the pattern of touch sensors in the multi-touch panel at which a touch event occurred can be viewed as an “image” of touch (e.g. a pattern of fingers touching the panel).) The capacitance between drive and sense electrodes can appear as a stray capacitance when the given row is held at direct current (DC) voltage levels and as a mutual signal capacitance Csig when the given row is stimulated with an alternating current (AC) signal. The presence of a finger or other object near or on the touch sensor panel can be detected by measuring changes to a signal charge Qsig present at the pixels being touched, which is a function of Csig. 
     Computing system  100  can also include host processor  128  for receiving outputs from processor subsystems  102  and performing actions based on the outputs that can include, but are not limited to, moving an object such as a cursor or pointer, scrolling or panning, adjusting control settings, opening a file or document, viewing a menu, making a selection, executing instructions, operating a peripheral device connected to the host device, answering a telephone call, placing a telephone call, terminating a telephone call, changing the volume or audio settings, storing information related to telephone communications such as addresses, frequently dialed numbers, received calls, missed calls, logging onto a computer or a computer network, permitting authorized individuals access to restricted areas of the computer or computer network, loading a user profile associated with a user&#39;s preferred arrangement of the computer desktop, permitting access to web content, launching a particular program, encrypting or decoding a message, and/or the like. Host processor  128  can also perform additional functions that may not be related to panel processing, and can be coupled to program storage  132  and display device  130  such as an LCD display for providing a UI to a user of the device. In some embodiments, host processor  128  can be a separate component from touch controller  106 , as shown. In other embodiments, host processor  128  can be included as part of touch controller  106 . In still other embodiments, the functions of host processor  128  can be performed by processor subsystem  102  and/or distributed among other components of touch controller  106 . Display device  130  together with touch sensor panel  124 , when located partially or entirely under the touch sensor panel, can form touch screen  118 . 
     Note that one or more of the functions described above can be performed, for example, by firmware stored in memory (e.g., one of the peripherals) and executed by processor subsystem  102 , or stored in program storage  132  and executed by host processor  128 . The firmware can also be stored and/or transported within any computer-readable storage medium for use by or in connection with an instruction execution system, apparatus, or device, such as a computer-based system, processor-containing system, or other system that can fetch the instructions from the instruction execution system, apparatus, or device and execute the instructions. In the context of this document, a “computer-readable storage medium” can be any medium that can contain or store the program for use by or in connection with the instruction execution system, apparatus, or device. The computer readable storage medium can include, but is not limited to, an electronic, magnetic, optical, electromagnetic, infrared, or semiconductor system, apparatus or device, a portable computer diskette (magnetic), a random access memory (RAM) (magnetic), a read-only memory (ROM) (magnetic), an erasable programmable read-only memory (EPROM) (magnetic), a portable optical disc such a CD, CD-R, CD-RW, DVD, DVD-R, or DVD-RW, or flash memory such as compact flash cards, secured digital cards, USB memory devices, memory sticks, and the like. 
       FIG. 2   a  illustrates exemplary mutual capacitance touch sensor panel  200  according to an embodiment of the present disclosure.  FIG. 2   a  indicates the presence of a stray capacitance Cstray at each pixel  202  located at the intersection of a row  204  and a column  206  trace (although Cstray for only one column is illustrated in  FIG. 2   a  for purposes of simplifying the figure). In the example of  FIG. 2   a , AC stimuli Vstim  214 , Vstim  215  and Vstim  217  can be applied to several rows, while other rows can be connected to DC. Vstim  214 , Vstim  215  and Vstim  217  can be at different frequencies and phases, as will be explained later. Each stimulation signal on a row can cause a charge Qsig=Csig×Vstim to be injected into the columns through the mutual capacitance present at the affected pixels. A change in the injected charge (Qsig_sense) can be detected when a finger, palm or other object is present at one or more of the affected pixels. Vstim signals  214 ,  215  and  217  can include one or more bursts of sine waves. Note that although  FIG. 2   a  illustrates rows  204  and columns  206  as being substantially perpendicular, they need not be so aligned, as described above. As described above, each column  206  can be connected to a sense channel (see sense channels  108  in  FIG. 1 ). 
       FIG. 2   b  is a side view of exemplary pixel  202  in a steady-state (no-touch) condition according to various embodiments. In  FIG. 2   b , an electric field of electric field lines  208  of the mutual capacitance between column  206  and row  204  traces or electrodes separated by dielectric  210  is shown. 
       FIG. 2   c  is a side view of exemplary pixel  202  in a dynamic (touch) condition. In  FIG. 2   c , finger  212  has been placed near pixel  202 . Finger  212  can be a low-impedance object at signal frequencies, and can have an AC capacitance Cfinger from the column trace  204  to the body. The body can have a self-capacitance to ground Cbody of about 200 pF, where Cbody is much larger than Cfinger. If finger  212  blocks some electric field lines  208  between the row and column electrodes (those fringing fields that exit the dielectric and pass through the air above the row electrode), those electric field lines can be shunted to ground through the capacitance path inherent in the finger and the body, and as a result, the steady state signal capacitance Csig can be reduced by ΔCsig. In other words, the combined body and finger capacitance can act to reduce Csig by an amount ΔCsig (which can also be referred to herein as Csig_sense), and can act as a shunt or dynamic return path to ground, blocking some of the electric fields as resulting in a reduced net signal capacitance. The signal capacitance at the pixel can become Csig−ΔCsig, where Csig represents the static (no touch) component and ΔCsig represents the dynamic (touch) component. Note that Csig−ΔCsig may always be nonzero due to the inability of a finger, palm or other object to block all electric fields, especially those electric fields that remain entirely within the dielectric material. In addition, it should be understood that as a finger is pushed harder or more completely onto the multi-touch panel, the finger can tend to flatten, blocking more and more of the electric fields, and thus ΔCsig can be variable and representative of how completely the finger is pushing down on the panel (i.e. a range from “no-touch” to “full-touch”). 
       FIG. 3  is a more detailed block diagram of an example touch controller  106  (e.g., a multi-touch controller) according to an embodiment of the present disclosure. Receive (RX) section  107  of touch controller  106  can include miscellaneous channels  305  (e.g., channels for infrared sensors, temperature sensors, etc.) and a total of N receive channels, such as sense channels  307 . Sense channels  307  can be connected to an offset compensator  309 . Multistage vector demodulation engine  109  can include a digital demodulation section  313 , a result memory  315 , and a vector operator  317 . Digital demodulation section  313  can be connected to a receive NCO  319 , and vector operator  317  can be connected to a decode matrix RAM  321  and connected to a result RAM  323 . Transmit (TX) section  114  can include a transmit logic  327 , a transmit DAC  329 , and a total of M transmit channels  333 . Transmit NCO  335  can provide a clock to transmit logic and TX DAC and charge pump  115  can provide power to the transmit channels. Transmit channels  333  can be connected to a stimulation matrix RAM  337  via an analog bus  339 . Decode matrix RAM  321 , result RAM  323 , and stimulation matrix RAM  337  could be, for example, part of RAM  112 . Processor subsystem  102  can store and update, for example, a decode matrix in decode matrix RAM  321  and a stimulation matrix in stimulation matrix RAM  337 , initialize the multi-touch subsystem, for example, process data from the receive channels and facilitate communications with the host processor. 
       FIG. 3  shows processor subsystem  102 , panel scan logic  110 , and host processor  128 .  FIG. 3  also shows a clock generator  343  and a processor interface  347 . Various components of touch controller  106  can be connected together via a peripheral bus  349 . Processor interface  347  can be connected to host processor  128  via a processor interface (PI) connection  353 . 
       FIG. 4  illustrates details of one of the sense channels  307  and digital demodulation section  313  according to an embodiment of the present disclosure. As shown in  FIG. 4 , sense channel  307  can include a transimpedance amplifier (TIA)  401 , an anti-alias filter (AAF)  403 , and an analog-to-digital converter (ADC)  405 . Digital demod section  313  can include a programmable delay  407 , a mixer (signal multiplier)  409 , and an integrator  411 . In each step of the scan, TIA  401  of sense channel  307  can receive a composite signal charge along with a programmable offset charge. 
     In some cases, the sense signal can be adjusted by offset compensator  309  prior to being input to TIA  401 . Adjusting the offset of the digital signal can reduce the dynamic range of some stimulation signals generated from highly variable stimulation matrices. In particular, some highly variable stimulation matrices may result in sense signals having a dynamic range greater than the dynamic input range of TIA  401 ; that is, the maximum signal magnitude that the amplifier can accept before the charge amplifier saturates. For example, in the case that the stimulation matrix is a Hadamard matrix, in one of the steps in the scan all of the channels are driven with stimulation signals having the same phase, and it is possible that all of the resulting component sense signals would add up to generate a composite sense signal with an amplitude that saturates TIA  401 . In this case, offset compensation would be used to subtract sufficient charge from the input charge as to prevent the charge amplifier from saturating. Offset compensation during a scan can be performed on-the-fly, that is, different offset compensation can be applied during different steps of the scan. 
     In another example embodiment, saturation of TIA  401  may be mitigated by adjusting, for example, the feedback of the amplifier. As is described herein, feedback of TIA  401  can be resistive, in addition to capacitive feedback, described in previous U.S. patent application Ser. No. 12/283,423, for example. In this case, individual sense channels can be adjusted, but the adjustment can remain the same for each step in a scan. This approach can be acceptable in the case that the stimulation matrix being used causes the same or similar imbalances of signals in the channels throughout the scan, and the amount of adjustment is not too great, e.g., up to a factor of 2. For example, using a circulant matrix as the stimulation matrix can cause a fixed imbalance across all steps. 
     The processing of a sense signal to obtain a value for Qsig_total is described below in reference to processing a single component of the sense signal of one sense channel (resulting from the stimulation of one of the channel&#39;s pixels) to obtain a single Qsig component of Qsig_total for that sense channel. However, it is understood that the analysis applies to all component signals, and that an actual Qsig_total result may be understood as simply a superposition of the individual Qsig results of the other component signals. 
     When a stimulation signal, Vstim, is applied to the drive line of a pixel, the AC portion of the stimulation signal, Vstim_AC(t), can be coupled through to the sense line, generating a signal charge Qsig(t) that tracks Vstim_AC(t) with an amplitude proportional to the signal capacitance Csig of the pixel. Qsig(t) can be expressed as:
 
 Qsig ( t )= Csig×Vstim   —   AC ( t )  (1)
 
A feedback capacitance, for example, in the feedback path of TIA  401  can convert the injected signal charge into an output voltage relative to the reference voltage of VREF of the charge amplifier
 
                       V   amp_out     ⁡     (   t   )       =       Qsig   ⁡     (   t   )         C   f               (   2   )               
Substituting for Qsig(t) using equation (1) yields:
 
                       V   amp_out     ⁡     (   t   )       =       Csig     C   f       ×   Vstim_AC   ⁢     (   t   )               (   3   )               
Thus, TIA  401  can output a signal whose amplitude is the stimulus amplitude Vamp_out(t) scaled by the gain (Csig/Cf) of the charge amplifier. In more general terms, sensor panel  124  can add an amplitude modulation to the drive signal, the amplitude modulation carrying information about something to be sensed, e.g. the a finger, etc.
 
     The output of TIA  401  can be fed into AAF  403 . AAF  403  can attenuate noise components above the nyquist sampling limit of the ADC sufficiently to prevent those components from aliasing back into the operating frequency range of touch controller  106 . Furthermore, AAF  403  can attenuate any noise outside the frequency operating range of touch controller  106  and therefore helps to improve the Signal-to-Noise ratio. It also can be important to properly select the sampling clock FCLK_DAC of the TX DAC. Generating a signal of frequency FSTM at the TX DAC clock rate can introduce images in the spectrum of the TX DAC output signal at n*FCLK_DAC+/−FSTM whereas N=1,2 . . . , to infinity. The images will appear in the composite signal entering the receive channel. Upon sampling the composite signal with the ADC in the receive channel, those images can be folded around the sampling frequency FCLK_ADC at which the ADC samples the composite touch signal. The output of the ADC therefore can have the following frequency components: N*(FCLK_DAC+/−FCLK_ADC)+/−FSTM. If the DAC and ADC clock rate FCLK_DAC and FCLK_ADC, respectively, are the same frequency, these images can appear in the pass-band. In the above example, one possible frequency component can be (FCLK_DAC−FCLK_ADC)+FSTM=FSTM and therefore can appear as an undesirable in-band component which can lead to reduced SNR and therefore reduced touch performance. Therefore, it can be beneficial to select a TX DAC sampling frequency FCLK_DAC that is different from the ADC sampling rate. This can prevent the images from folding back into the pass-band. In one embodiment, FCLK_DAC can be twice of the ADC clock rate FCLK_ADC. The two clock sources should be correlated, i.e. based on the same master clock. It can be beneficial to make the DAC sampling clock higher in frequency than the ADC sampling clock as DACs can consume less power than the power consumed by all ADCs combined for the same increase in sampling clock frequency. 
     The output of AAF  403  can be converted by ADC  405  into a digital signal, which can be sent from sense channel  307  to digital demodulation section  313 . Digital demodulation section  313  can demodulate the digital signal received from sense channel  307  using a homodyne mixing process in which the signal is multiplied with a demodulation signal of the same frequency. In order to increase the efficiency of the mixing process, it may be desirable to adjust the phase of the sense channel output signal to match the phase of the demodulation signal. Stimulating a pixel of sensor panel  124  with Vstim+ and processing the resulting sense signal as described above can result in the following output from sense channel  307 : 
     
       
         
           
             
               
                 
                   
                     
                       V 
                       
                         
                           sense_ch 
                           ⁢ 
                           _outV 
                         
                         + 
                       
                     
                     ⁡ 
                     
                       ( 
                       t 
                       ) 
                     
                   
                   = 
                   
                     
                       Csig 
                       
                         C 
                         f 
                       
                     
                     × 
                     
                       V 
                       0 
                     
                     ⁢ 
                     
                       sin 
                       ⁡ 
                       
                         ( 
                         
                           
                             ω 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             t 
                           
                           + 
                           θ 
                         
                         ) 
                       
                     
                   
                 
               
               
                 
                   ( 
                   4 
                   ) 
                 
               
             
           
         
       
         
         
           
             where: V 0 =the amplitude of the AC portion of Vstim=2.25V
           θ=the relative phase delay between the signal output of ADC  405  and the demodulation signal for a given sense channel
 
For stimulation with Vstim−, the resulting output from ADC  405  can be:
   
         
           
         
       
    
                       V       sense_ch   ⁢   _outV     -       ⁡     (   t   )       =       Csig     C   f       ×     V   0     ⁢     sin   ⁡     (       ω   ⁢           ⁢   t     +     180   ⁢   °     +   θ     )                 (   5   )               
The relative phase delay θ can be an aggregate of delays caused by various elements of the system, such as the geometry of the signal paths, the operation of the output buffers, etc. In general, the various delays in the system can be separated into two categories, delays that apply equally to all drive lines of a sense channel, referred to as global delays herein, and delays that vary among the drive lines of the sense channel, referred to as individual line delays herein. In other words, global delays can affect all component signals of the composite sense signal equally, while individual line delays can result in different amounts of delay for different component signals. The relative phase delay can be represented as:
 
θ= DCL +φ( R )  (6)
         where: DCL=the sum of all global delays (referred to herein as the composite global delay) affecting a sense channel
           φ(R)=the individual line delay associated with drive line R of a sense channel
 
Substituting equation (6) into equations (4) and (5) yields:
   
               

                       V       sense_ch   ⁢   _outV     +       ⁡     (   t   )       =       Csig     C   f       ×     V   0     ⁢     sin   ⁡     (       ω   ⁢           ⁢   t     +     D   ⁢           ⁢   C   ⁢           ⁢   L     +     ϕ   ⁡     (   R   )         )                 (   7   )                   V       sense_ch   ⁢   _outV     -       ⁡     (   t   )       =       Csig     C   f       ×     V   0     ⁢     sin   ⁡     (       ω   ⁢           ⁢   t     +     180   ⁢   °     +     D   ⁢           ⁢   C   ⁢           ⁢   L     +     ϕ   ⁡     (   R   )         )                 (   8   )               
Since the global delays can affect all of the component signals of the sense signal equally, once the composite global delay DCL has been determined for a channel, the global portion of the phase delay of sense channel output signal can be removed by programmable delay  407 , yielding:
 
                         V     mixer_inV   +       ⁡     (   t   )       =       Csig     C   f       ×     V   0     ⁢     sin   ⁡     (       ω   ⁢           ⁢   t     +     ϕ   ⁡     (   R   )         )           ⁢     
     ⁢         V     mixer_inV   -       ⁡     (   t   )       =       Csig     C   f       ×     V   0     ⁢     sin   ⁡     (       ω   ⁢           ⁢   t     +     180   ⁢   °     +     ϕ   ⁡     (   R   )         )                   (   9   )               
as the signals corresponding to Vstim+ and Vstim−, respectively, that are input into mixer  409 .
 
     Since the individual line delays can be different for different signal components of the sense signal, the individual line delays cannot be removed from the sense signal simply by using a single phase adjustment to the composite sense signal, such as the phase adjustment made by programmable delay  407 . However, the individual line delays may be accounted for by the compensated phase matrix {tilde over (M)} comp   −1 , which is described in more detail below. 
     The phase-adjusted signal can be sent from programmable delay  407  to mixer  409 . Mixer  409  can multiply the phase-adjusted signal with a demodulation signal,
 
 V   demod =sin(ω t ),  (11)
 
which can be generated by RX NCO  319  based on a master oscillator  415 . It is noted that the mixing can be performed using digital signals. This can provide higher resolution than in some previous designs, which can result in improved suppression of noise.
 
     The resulting demodulated signal output from mixer  409  can be expressed as: 
                       V     mixer_outV   +       ⁡     (   t   )       =       1   2     ×     Csig     C   f       ×     V   0     ×     (       cos   ⁡     (     ϕ   ⁡     (   R   )       )       -     cos   ⁡     (       2   ⁢   ω   ⁢           ⁢   t     +     ϕ   ⁡     (   R   )         )         )               (   12   )                   V     mixer_outV   -       ⁡     (   t   )       =       1   2     ×     Csig     C   f       ×     V   0     ×     (       cos   ⁡     (       180   ⁢   °     +     ϕ   ⁡     (   R   )         )       -     cos   ⁡     (       2   ⁢   ω   ⁢           ⁢   t     +     180   ⁢   °     +     ϕ   ⁡     (   R   )         )         )               (   13   )               
The mixer output can be integrated by integrator  411 , yielding:
 
                     V     int_outV   +       =       1   2     ×     Csig     C   f       ×     V   0     ×     cos   ⁡     (     ϕ   ⁡     (   R   )       )                 (   14   )                 V     int_outV   -       =       1   2     ×     Csig     C   f       ×     V   0     ×     cos   ⁡     (       180   ⁢   °     +     ϕ   ⁡     (   R   )         )                 (   15   )               
Since the integrator has essentially a low pass response, the high frequency component cos (2ωt+180°+φ(R)) can be eliminated leaving only the DC component.
 
Scaling of the results in integrator  411  by a factor of 2C f  results in output signals:
 
 V   int     —     scaledV+   =V   0 ×cos(φ( R ))× Csig , if  Vstim ( R )= Vstim+   (16)
 
 V   int     —     scaledV−   =V   0 ×cos(180°+φ( R ))× Csig , if  Vstim ( R )= Vstim−   (17)
 
from integrator  411 . In each step S in a scan of sensor panel  124 , drive lines  204  are driven with either Vstim+ or Vstim− drive signals based on the MUX_SEL values in stim matrix  407  for that step, each stimulation signal generating a component output (16) or (17) of integrator  411  for each sense channel. Thus, for a channel C, the output of integrator  411  can be a linear combination of corresponding components (16) and (17):
 
                         V     int_scaled   ⁢   _tot   ⁢   _C       ⁡     (   S   )       =         V   0     ×       W   C     ⁡     (     0   ,   S     )       ×     Csig   ⁡     (   0   )         +       V   0     ×       W   C     ⁡     (     1   ,   S     )       ×     Csig   ⁡     (   1   )         +     …   ⁢           ⁢     V   0     ×     (       M   ⁢     -     ⁢   1     ,   S     )     ×     Csig   ⁡     (     M   ⁢     -     ⁢   1     )             ⁢     
     ⁢       where   ⁢     :     ⁢           ⁢       W   C     ⁡     (     R   ,   S     )         =     {               cos   ⁡     (       ϕ   C     ⁡     (   R   )       )       ⁢     ⟶   if     ⁢     Vstim   ⁡     (     R   ,   S     )         =     Vstim   +                     cos   ⁡     (       180   ⁢   °     +       ϕ   C     ⁡     (   R   )         )       ⁢     ⟶   if     ⁢     Vstim   ⁡     (     R   ,   S     )         =     Vstim   -                         (   18   )               
The right hand side of equation (18) includes V 0  equal to the amplitude, Vstim, of the stimulation signals and W C (R,S) equal to the components of the compensated phase matrix {tilde over (M)} C     —     comp . Therefore, the output voltage of integrator  411 , V int     —     scaled     —     tot     —     C (S), at each step is simply the composite signal charge Qsig_tot C (S).
 
     The Qsig_tot C  values output by a channel&#39;s integrator  411  can be posted to result memory  315 , forming a Qsig_tot C  vector: 
                       Qsig_tot   C     ⁢     (   S   )       =     [             Qsig_tot   C     ⁢     (   0   )                   Qsig_tot   C     ⁢     (   1   )                   Qsig_tot   C     ⁢     (   2   )                                   Qsig_tot   C     ⁢     (     P   -   1     )             ]             (   19   )               
that is used in a decoding operation to determine the Csig values for that channel. An example vector decode operation according to various embodiments will now be described. Referring to  FIG. 3 , vector operator  317  can read the Qsig_tot C  vector from memory  315  and reads the decode matrix
 
                 M   ~     C_comp     -   1       Vstim         
from decode matrix RAM  321 . Vector operator  317  then performs vector multiplication of the Qsig_tot C  vector and the decode matrix
 
                 M   ~     C_comp     -   1       Vstim         
according to equation (9) to obtain the Csig C  vector for channel C:
 
                       C   ~     ⁢     sig   C       =     [             Csig   C     ⁡     (   0   )                   Csig   C     ⁡     (   1   )                   Csig   C     ⁡     (   2   )                                   Csig   C     ⁡     (     M   -   1     )             ]             (   20   )               
The Csig C  vector can be posted to result RAM  323 , where it can be read by other systems, such as processor subsystem  102 , host processor  128 , etc., for sensing touch by comparing the Csig C  vector components with known, static (no touch) values for Csig, for example.
 
     However, as an alternative to capacitive feedback in TIA  401 , resistive feedback in TIA  401  can be implemented.  FIG. 5  illustrates an exemplary single-ended analog front end (AFE), with an ADC  405  and bandpass filter (BPF)  500 . Of course, various ADCs may be employed (e.g., a 9-bit successive-approximation-register (SAR)) without departing from the scope of the present disclosure. Stage  1  of the AFE includes TIA  401 , which is described in greater detail below with respect to  FIG. 7 . 
     The output of TIA  401 , according to certain embodiments, can be input to BPF  500 , at Stage  2  of the AFE. The exemplary BPF  500  can provide 10 dB/octave of attenuation toward lower frequencies (i.e., below FSTM). As shown in  FIG. 5 , the output of the BPF  500  can be input to AAF  403 , at Stage  3  of the AFE, the output of which can be input to ADC  405 . 
     According to one embodiment, AAF  403  and ADC  405  can be replaced with a Sigma-Delta ADC  600 , for example, as shown in  FIG. 6 . Sigma-Delta ADC  600  can have low-pass filter response, which substantially provides the functionality of AAF  403 . AAF  403  is described below in greater detail with reference to  FIG. 9 . 
       FIG. 7  illustrates TIA  401 , of the AFE, with a feedback resistor RFB, according to an embodiment. As shown in  FIG. 7 , feedback resistor RFB can be coupled to the negative input of TIA  401  and to the output of TIA  401 . As resistors have a relatively small footprint, feedback resistor RFB can consume significantly less DIE real estate as compared to a capacitor, for example. 
     The single pole open loop transfer function of the amplifier can be: 
                     V   OUT     =         G   0       1   +     s     ω   0           ·     (       V   +     +     V   -       )               (   21   )               
V OUT =output voltage of the amplifier, ω 0  is the amplifier&#39;s pole frequency, G 0  is the DC gain bandwidth of the amplifier, V +  the voltage at the non-inverting input of the amplifier (=0V) and V_=the voltage at the inverting input;
 
The voltage into the inverting node of the preamplifier and hence across the stray capacitor is:
 
                     V   -     =         s   ·     C   SIG     ·     (       V   STM     -     V   -       )       +       (       V   OUT     -     V   -       )       R   FB           s   ·     C   S                 (   22   )               
After combining equations (21) ad (22) and further simplification this leads to:
 
                     V   OUT     =       -       s   ·     R   FB     ·     C   SIG         1   +         s   ·     R   FB     ·     (     1   +     s     ω   0         )       ⁢     (       C   SIG     +     C   S       )         G   0             ·     V   STM               (   23   )               
Therefore, the TIA has bandpass filter response. The gain bandwidth of the amplifier is chosen such that at the stimulus frequency range of interest the denominator of equation (23) is approximately 1 and equation (23) reduces to:
 
 H   TIA ( s )= s·R   FB   ·C   SIG   (24)
 
Equation (24) leads to:
 
 H   TIA (ω STM )=ω STM   ·R   FB   ·C   SIG   (25)
 
     Therefore for the stimulus frequency range of interest, TIA has high-pass filter response and the signal at the output of the preamplifier is phase-shifted by 90 degrees relative to VSTM. Note that the stray capacitance CS drops out of the equation as desired for the stimulus frequency range of interest. 
     RFB can be adjustable as to maximize the dynamic output range of VOUT of the TIA for a given stimulus frequency FSTM. It may be desirable to move the stimulus frequency as far above an interferer as possible. For example, a switching power supply can introduce noise into the touch subsystem at 100 Khz. In this scenario it can be beneficial to move FSTM to a higher frequency. A stimulus frequency of 200, 400 and 800 Khz can cause attenuation of the 100 Khz noise component in the TIA by 6, 12 and 18 dB, respectively. 
     The lowpass filter response above of the TIA  401  can add benefits as it can attenuate high frequency noise that can be induced into the touch mechanism and also can help to meet the nyquist attenuation requirements 
     According to an embodiment, at least one optional multiplexor  700  can be added at the input of TIA  401 , thereby allowing two or more inputs to be multiplexed into one dedicated channel, for example (e.g., 0.5 channels per input), thereby further reducing the required real estate of the DIE. 
       FIG. 8  is an illustration of an exemplary BPF  500 , according to an embodiment. As described above, BPF  500  can provide 10 dB/octave of attenuation at frequencies outside the passband. The gain of the band-pass filter allows adjustment of the available TIA output voltage range for noise. According to certain embodiments, the dynamic range at the output of the TIA or input of the bandpass filter can be: VSIG_DYN=VOUT_TOT/G_BPF, 
     Where: 
     VSIG_DYN is the dynamic range of the touch signal; 
     VTOT_DYN=total dynamic output range available for in-band signal component and noise (e.g. 0.9V); and 
     G_BPF=bandpass gain (e.g., 6). 
     In the above example, the dynamic range for the in-band signal component would be 0.9V/6=0.15V. 
     Therefore the remainder of the output range VTOT_DYN−VSIG_DYN=0.9V−0.15V=0.75V would be available for out of band noise components. The max output of TIA  401  can be 0.15V, therefore resulting in a 0.9 peak-to-peak voltage, for example, with a passband gain of 6 at BPF  500 . Thus, the headroom (0.9 V−0.15 V) can be left for external noise sources. Using this input voltage can remove a significant amount of interferers (above 0.15 V), while leaving the signal (i.e., the in-band component at or below 0.15 V). Input resistance R 1   a  and/or R 1   b  can be expressed as:
 
 R 1 =Q /( G* 2*π *f*C )  (26)
 
G (passband gain) and f (center frequency) are defined below, and Q is expressed as:
 
 Q= 0.5*( R 3*( R 1+( R 2/ 2 ))/( R 1*( R 2/ 2 ))){circumflex over ( 0 )}0.5  (27)
 
Q determines the bandwidth of the filter, such that the bandwidth equals f/Q (e.g., Q may be set to 2.5 according to an embodiment).
 
Attenuator Resistance R 2  can be Expressed as:
 
 R 2= 2 * Q /((2* Q 2 −G )*2*π* f*C )  (28)
 
(the  2  is for full differential inputs, but the  2  can be omitted for single ended inputs).
 
Feedback Resistance R 3  can be Expressed as:
 
 R 3 =Q /(π *f*C )  (29)
 
Passband Gain G can be Expressed as:
 
 G= 1/(( R 1/ R 3)*2)= R 3/(2* R 1)  (30)
 
Center Frequency f can be programmable using a 5-bit adjustment, for example, and can be expressed as:
 
 f =(1/(2*π *C ))*√(( R 1+( R 2/ 2 ))/( R 1*( R 2/ 2 )* R 3))  (31)
 
It is noted that this type of adjustment can yield a constant Q independent of the value of C.
 
       FIG. 9  illustrates an exemplary AAF  403  in an AFE, according to an embodiment. As described above with respect to  FIG. 4 , the output of TIA  401  can be fed into AAF  403 . AAF  403  can attenuate noise components above the nyquist sampling limit of ADC  405  sufficiently to prevent those components from aliasing back into the operating frequency range of touch controller  106 . Furthermore, AAF  403  can attenuate any noise outside the frequency operating range of touch controller  106  and therefore helps to improve the signal-to-noise ratio. The output of AAF  403  can be converted by ADC  405  into a digital signal, which can be sent from sense channel  307  to digital demodulation section  313 . Digital demodulation section  313  demodulates the digital signal received from sense channel  307  using a homodyne mixing process in which the signal is multiplied with a demodulation signal of the same frequency. Of course, in an alternate embodiment depicted in  FIG. 6 , AAF  403  can be left out when using a sigma-delta ADC  600 , for example. The sigma-delta ADC  600  can consume more power than ADC  405 , but can provide enhanced noise attenuation. 
       FIG. 10  illustrates various frequency response plots, according to embodiments. Referring to  FIG. 10 , the combined frequency response HRX of the receive (RX) channel can be expressed as:
 
 H   RX (ω)= H   TIA (ω)· H   BPF (ω)· H   AAF (ω)· H   ADC (ω)  (32)
 
Here H TIA (ω), H BPF (ω), H AAF (ω) and H ADC (ω) are the frequency transfer functions of the TIA, BPF, AAF and ADC, respectively. Plot  1010  shows an exemplary frequency response of TIA  401 . The lowpass filter response below the peak  1014  of the TIA  401  can be approximately 6 dB per octave or 20 dB per decade. RFB can be adjusted such that the in-band signal component H_TIA at the stimulus frequency FSTM as indicated by point  1012  is at least a factor of G_BPF=6 or 15.56 dB below the full-scale voltage VTOT_DYN at the output of TIA  401 .
 
     For the given topology, the peak of the transfer function plot  1010  as indicated by point  1014  can be set by the feedback network of the TIA  401  and can be typically well above point  1012 . Therefore, the TIA  401  can operate as a true differentiator at frequencies below FSTM, which may be desirable. Generally, it may be desirable to move point  1014  as close to point  1012  as possible in order to prevent noise frequencies above point  1012  to be amplified due to the high-pass filter response of the TIA  401   
     Plot  1020  shows an exemplary frequency response of the BPF  500 . Point  1024  is indicative of the center frequency FC of the BPF  500  and can be tuned to a given stimulus frequency as to maximize the in-band signal component while suppressing noise components outside a given pass-band. Points  1023  and  1025  can be the −3 dB points and the difference in frequency between points  1023  and  1024  can represent the bandwidth of the BPF which is equivalent to the center frequency of the BPF  500  divided by its quality factor Q_BPF. 
     Plot  1030  shows an exemplary frequency response of the AAF  403 . The AAF  403  may be an active or passive filter dependent on the anti-aliasing requirements. If a 2 nd  or higher order frequency response is desired it becomes necessary to use an active filter topology, which may yield a frequency response similar to that shown in plot  1030 . 
     Point  1032  indicates the center or resonant frequency of the AAF  403  and  1032  the −3 dB point. The range below point  1032  is the passband with the gain being ideally 1 (or 0 dB). For a AAF  403  of 2 nd  order the roll-off  1033  may be up to close to 40 dB/decade. 
     Plot  1040  shows an exemplary frequency response of the entire receive (RX) channel. If, for example, the low-pass filter roll-off of the TIA  401  is 20 dB/decade and the BPF  500  lowpass filter roll-off is 30 dB per decade and the AAF  403  can have a gain of 0 dB/decade, the combined roll-off of the entire RX channel can be 50 dB per decade, as indicated by Eq. (34). 
     The frequency response H_RX between points  1012  and  1014  can be that of a low-pass filter. However, the highpass filter response  1012  of the TIA  401  up to that point counter can act the low-pass filter response of the BPF  500 . Therefore, the combined roll-off of the TIA  401  and BPF  500  can be 30 dB/decade−20 dB/decade=10 dB/decade. 
     The AAF  403  compensates for the high-pass filter response of the TIA  401  and can be necessary to meet the anti-aliasing requirements of the ADC  405 , for example. If, for example, the AAF  403  has a lowpass filter roll-off of 35 dB, the total low-pass filter roll-off in segment  1042  can be 10 dB/decade+35 dB/decade=45 dB/decade. 
     In segment  1043 , the frequency response of the TIA  401  can be past point  1014  and therefore the combined low-pass filter roll-off can be the sum of the roll-offs of segments  1015 ,  1026  and  1033  for a total of 20 dB/decade+30 dB/decade+35 dB/decade=85 dB/decade 
     The AAF  403  can be provided to meet the overall attenuation requirements in the aliased frequency range FSAMP-FSTM(MAX) to FSAMP-FSTM(MIN). Also, the AAF  403  can compensate for the high-pass filter response of the TIA  401  above the selected RX center frequency. It can be beneficial to make the AAF  403  cutoff frequency programmable so it can track the BPF  500  center frequency in order to optimize the lowpass filter roll-off and compensate for the high pass filter response of the 
     Dependent on the type of ADC architecture selected, the AAF  403  may not be needed. The AAF  403  can be a second order active filter with programmable cut-off frequency. The transfer function of the AAF  403  filter can be provided as: 
                     H   ⁡     (   s   )       =       GAIN_AAF   ·     ω   0   2           s   2     +         ω   0     Q_AAF     ·   s     +     ω   0   2                 (   33   )               
Where: GAIN_AAF=gain of AAF
 
                   GAIN_AAF   =     -       R   FB_AAF       R   IN_AAF                 (   34   )               
Q_AAF=quality factor of AAF:
 
     
       
         
           
             
               
                 
                   
                     Q_AAF 
                     = 
                     
                       1 
                       
                         
                           ω 
                           0 
                         
                         · 
                         
                           C 
                           FBK_AAF 
                         
                         · 
                         
                           
                             
                               R 
                               IN_AAF 
                             
                             + 
                             
                               2 
                               · 
                               
                                 R 
                                 
                                   IN_AAF 
                                   ⁢ 
                                   
                                       
                                   
                                   ⁢ 
                                   2 
                                 
                               
                             
                           
                         
                       
                     
                   
                   ⁢ 
                   
                     
 
                   
                   ⁢ 
                   
                     
                       ω 
                       0 
                     
                     = 
                     
                       
                         “ 
                         resonant 
                         ” 
                       
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       frequency 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       of 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       A 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       A 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       F 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       
                         ( 
                         
                           in 
                           ⁢ 
                           
                               
                           
                           ⁢ 
                           radians 
                         
                         ) 
                       
                       ⁢ 
                       
                         : 
                       
                     
                   
                 
               
               
                 
                   ( 
                   35 
                   ) 
                 
               
             
             
               
                 
                   
                     ω 
                     0 
                   
                   = 
                   
                     1 
                     
                       
                         
                           C 
                           IN_AAF 
                         
                         · 
                         
                           C 
                           FBK_AAF 
                         
                         · 
                         
                           R 
                           
                             IN_AAF 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             1 
                           
                         
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                           R 
                           
                             IN_AAF 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             2 
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   36 
                   ) 
                 
               
             
           
         
       
     
     Due to the peaking of the AAF  403 , the actual cut-off frequency FC_AAF can be calculated according to the following formula: 
                   FC_AAF   =       F   ⁢           ⁢   S   ⁢           ⁢     F   ·     ω   0           2   ·   π               (   37   )               
Where: FSF=Frequency scaling factor.
 
     The cutoff frequency can be adjusted by making C IN     —     AAF  and C FBK     —     AAF  programmable. 
       FIG. 11  illustrates an exemplary analog front end, according to various embodiments. This implementation combines a TIA, bandpass and ADC functions into one single block. The resulting device  1100  can be essentially a sigma delta convertor with bandpass filter response. Summer  1105  sums the sense signal  1101  with the feedback signal  1102  from ADC  1130 . The summing node  1105  can be held at virtual ground in order to reject the stray capacitance CS of the sense line. Summing node  1105  can be the non-inverting input node of a TIA, and may nor may not be part of the bandpass filter  1110 . The resulting signal  1103  can be fed into a bandpass filter which can have a programmable center frequency. The bandpass filtered signal  1111  can be fed into a quantizer  1115 , which can have one or multiple quantization levels. The digitized signal  1122  can then be converted to an analog signal  1102  and fed back to summing node  1105 . The quantization noise VNZ introduced by quantizer  1115  can be modeled by summer  1120 . Gain adjustment can be accomplished via digital gain adjustment  1131  which scales the feedback signal  1102  therefore allowing gain adjustment. The digitized signal can also be fed into Decimator  1125  and the decimated signal  1126  can be then passed on for further processing. Note that the sigma delta modulator described can be a higher order sigma delta modulator and can have tunable resonators as bandpass filters. Similar sigma delta modulators are described in U.S. Pat. No. 6,218,972 entitled “Tunable bandpass sigma-delta digital receiver” and U.S. Pat. No. 5,027,120 entitled “Delta-sigma convertor with bandpass filter for noise reduction in receivers”. 
     As a potential advantage, according to the embodiment depicted in  FIG. 11 , noise may be rejected at the input (Stage 1) of AFE while the ADC&#39;s  600  dynamic output range is maximized, and signal-to-noise ratio can be boosted. This provides a single-stage approach, with no requirement for a separate AAF  403 . However, power consumption may be higher, as compared to using a biquad filter, as described below with reference to  FIG. 12 . 
     As yet another alternative, BPF  500  may be implemented in Stage 1 with the AFE including a biquad bandpass filter implementing nested feedback as shown in  FIG. 12 . In this case, the biquad bandpass filter includes a virtual ground capability. As shown in  FIG. 12 , the biquad filter can be based on a two state variable filter  1240 , which can be comprised of two integrators  1210  and  1220  and a summing node. 
       FIG. 13  shows a circuit example of a Tow Thomas biquad filter, which may also be implemented at Stage  1 . Here CSIG replaces the input resistor R 1  commonly used in the Tow Thomas biquad filter topology. Invertor  1350  can be utilized to establish proper polarity and is not needed in fully differential implementations. The inverting input of the first integrator  1310  can serve as the summing node. The output of first integrator  1310  can be fed into second integrator  1320 . Since the summing node can be held at virtual ground, it can reject CS, the stray capacitance imposed on the input of the biquad filter by the sensor panel. With the bandpass filtering at the input (Stage  1 ) of the AFE, noise can be rejected at Stage  1 , while maximizing the dynamic output range and boosting the signal-to-noise ratio. However, a biquad filter may consume more DIE real estate, as compared to the resistive feedback TIA  401  using only feedback resistor RFB, as shown in  FIG. 7 . Various combinations of capacitances, resistances and Q may be programmable. 
     Thereafter, Stage  2  can include sigma-delta ADC  600 , for example. Thus, no separate anti-aliasing filter would be required, due to the combined lowpass filter response of the biquad filter and the sigma delta convertor, and the relative high oversampling rate of the sigma delta convertor. 
       FIG. 14   a  illustrates an example mobile telephone  1436  that can include touch sensor panel  1424  and display device  1430 , the touch sensor panel including an analog front end design according to one of the various embodiments described herein. 
       FIG. 14   b  illustrates an example digital media player  1440  that can include touch sensor panel  1424  and display device  1430 , the touch sensor panel including an analog front end design according to one of the various embodiments described herein. 
       FIG. 14   c  illustrates an example personal computer  1444  that can include touch sensor panel (trackpad)  1424  and display  1430 , the touch sensor panel and/or display of the personal computer (in embodiments where the display is part of a touch screen) including an analog front end design according to the various embodiments described herein. 
     While various embodiments have been described above, it should be understood that they have been presented by way of example only, and not by way of limitation. Likewise, the various diagrams may depict an example architectural or other configuration for the disclosure, which is done to aid in understanding the features and functionality that can be included in the disclosure. The disclosure is not restricted to the illustrated example architectures or configurations, but can be implemented using a variety of alternative architectures and configurations. Additionally, although the disclosure is described above in terms of various exemplary embodiments and implementations, it should be understood that the various features and functionality described in one or more of the individual embodiments are not limited in their applicability to the particular embodiment with which they are described. They instead can be applied alone or in some combination, to one or more of the other embodiments of the disclosure, whether or not such embodiments are described, and whether or not such features are presented as being a part of a described embodiment. Thus the breadth and scope of the present disclosure should not be limited by any of the above-described exemplary embodiments.

Metadata:
Filing Date: 20090911
Publication Date: 20111004
Grant Date: 20111004
Priority Date: 20090911
Inventors: HOTELLING STEVEN PORTER
KRAH CHRISTOPH HORST
Assignee: APPLE INC
CPC Classifications: [{"code": "H03M3/494", "inventive": true, "first": true, "tree": "[]"}, {"code": "G06F3/0446", "inventive": true, "first": false, "tree": "[]"}, {"code": "G06F3/04182", "inventive": true, "first": false, "tree": "[]"}, {"code": "H03M3/494", "inventive": true, "first": true, "tree": "[]"}]
Family ID: 43729979