Query: i. High Efficiency RF and Microwave Solid Sate Power Amplifiers

Candidate papers:

abstract: Preface. About the Authors. Acknowledgments. 1 Power Amplifier Fundamentals. 1.1 Introduction. 1.2 Definition of Power Amplifier Parameters. 1.3 Distortion Parameters. 1.4 Power Match Condition. 1.5 Class of Operation. 1.6 Overview of Semiconductors for PAs. 1.7 Devices for PA. 1.8 Appendix: Demonstration of Useful Relationships. 1.9 References. 2 Power Amplifier Design. 2.1 Introduction. 2.2 Design Flow. 2.3 Simplified Approaches. 2.4 The Tuned Load Amplifier. 2.5 Sample Design of a Tuned Load PA. 2.6 References. 3 Nonlinear Analysis for Power Amplifiers. 3.1 Introduction. 3.2 Linear vs. Nonlinear Circuits. 3.3 Time Domain Integration. 3.4 Example. 3.5 Solution by Series Expansion. 3.6 The Volterra Series. 3.7 The Fourier Series. 3.8 The Harmonic Balance. 3.9 Envelope Analysis. 3.10 Spectral Balance. 3.11 Large Signal Stability Issue. 3.12 References. 4 Load Pull. 4.1 Introduction. 4.2 Passive Source/Load Pull Measurement Systems. 4.3 Active Source/Load Pull Measurement Systems. 4.4 Measurement Test sets. 4.5 Advanced Load Pull Measurements. 4.6 Source/Load Pull Characterization. 4.7 Determination of Optimum Load Condition. 4.8 Appendix: Construction of Simplified Load Pull Contours through Linear Simulations. 4.9 References. 5 High Efficiency PA Design Theory. 5.1 Introduction. 5.2 Power Balance in a PA. 5.3 Ideal Approaches. 5.4 High Frequency Harmonic Tuning Approaches. 5.5 High Frequency Third Harmonic Tuned (Class F) 5.6 High Frequency Second Harmonic Tuned. 5.7 High Frequency Second and Third Harmonic Tuned. 5.8 Design by Harmonic Tuning. 5.9 Final Remarks. 5.10 References. 6 Switched Amplifiers. 6.1 Introduction. 6.2 The Ideal Class E Amplifier. 6.3 Class E Behavioural Analysis. 6.4 Low Frequency Class E Amplifier Design. 6.5 Class E Amplifier Design with 50# Duty cycle. 6.6 Examples of High Frequency Class E Amplifiers. 6.7 Class E vs. Harmonic Tuned. 6.8 Class E Final Remarks. 6.9 Appendix: Demonstration of Useful Relationships. 6.10 References. 7 High Frequency Class F Power Amplifiers. 7.1 Introduction. 7.2 Class F Description Based on Voltage Wave shaping. 7.3 High Frequency Class F Amplifiers. 7.4 Bias Level Selection. 7.5 Class F Output Matching Network Design. 7.6 Class F Design Examples. 7.7 References. 8 High Frequency Harmonic Tuned Power Amplifiers. 8.1 Introduction. 8.2 Theory of Harmonic Tuned PA Design. 8.3 Input Device Nonlinear Phenomena: Theoretical Analysis. 8.4 Input Device Nonlinear Phenomena: Experimental Results. 8.5 Output Device Nonlinear Phenomena. 8.6 Design of a Second HT Power Amplifier. 8.7 Design of a Second and Third HT Power Amplifier. 8.8 Example of 2nd HT GaN PA. 8.9 Final Remarks. 8.10 References. 9 High Linearity in Efficient Power Amplifiers. 9.1 Introduction. 9.2 Systems Classification. 9.3 Linearity Issue. 9.4 Bias Point Influence on IMD. 9.5 Harmonic Loading Effects on IMD. 9.6 Appendix: Volterra Analysis Example. 9.7 References. 10 Power Combining. 10.1 Introduction. 10.2 Device Scaling Properties. 10.3 Power Budget. 10.4 Power Combiner Classification. 10.5 The T junction Power Divider. 10.6 Wilkinson Combiner. 10.7 The Quadrature (90 Hybrid. 10.8 The 180 Hybrid (Ring Coupler or Rat race) 10.9 Bus bar Combiner. 10.10 Other Planar Combiners. 10.11 Corporate Combiners. 10.12 Resonating Planar Combiners. 10.13 Graceful Degradation. 10.14 Matching Properties of Combined PAs. 10.15 Unbalance Issue in Hybrid Combiners. 10.16 Appendix: Basic Properties of Three port Networks. 10.17 References. 11 The Doherty Power Amplifier. 11.1 Introduction. 11.2 Doherty's Idea. 11.3 The Classical Doherty Configuration. 11.4 The 'AB C' Doherty Amplifier Analysis. 11.5 Power Splitter Sizing. 11.6 Evaluation of the Gain in a Doherty Amplifier. 11.7 Design Example. 11.8 Advanced Solutions. 11.9 References. Index.
author_names: Paolo Colantonio

Franco Giannini

Ernesto Limiti
title: High Efficiency RF and Microwave Solid State Power Amplifiers
year: 2009

abstract: In the last 10 years or so wide bandgap transistors (SiC MESFETs and GaN HEMTs) have appeared on the market for high power RF/microwave transistors. They offer higher power density and higher voltage operation, which in turn are associated with much lower parasitic capacitances and much higher load line dynamic resistance, and hence wider bandwidth applications. Of the two kinds the GaN HEMTs offer higher gain performance and became dominant on the market. The much wider bandwidth matching networks though could not be designed optimally with the traditional Smith Chart and optimization techniques [1] The new requirements for broadband high power and high efficiency performance requires new more sophisticated matching networks synthesis techniques such as the real frequency technique [4] [7]
author_names: Ivan Boshnakov

Anna Wood

Simon Taylor
title: RF and microwave solid state power amplifiers design requires specialised engineering
year: 2012

abstract: Energy efficient conversion of electrical grid power into radio frequency power is becoming one of the key aspects of future accelerators. A significant part of the initial investment and running costs of these machines will be determined by the cost and efficiency of their RF systems. For largescale accelerators, which are proposed for instance by the International Linear Collider Study or by the Future Circular Collider Study, RF efficiency will likely be a determining factor in the approval process. Efforts are already in place to stretch the efficiency of existing RF sources to higher levels. Well known technologies such as klystrons are being reinvented with modern day beam physics, inductive output tubes are being combined for higher power output, and modular solid state amplifiers are becoming more popular as RF sources for accelerators. This talk will give an overview of recent advances and trends in RF source developments. It will discuss future needs and the strategy towards higher efficiency devices for the benefit of the accelerator community. THE NEED FOR HIGHER EFFICIENCY RF SOURCES The electrical power consumption of future accelerators will be driven to a large part by their RF systems. This is particularly true for electron colliders, circular (e.g. FCC) or linear (e.g. ILC or CLIC) which need RF systems in the order of 100 MW or above (see Table 1, [1 6] Other examples with significant RF power consumption are high power hadron linacs (e.g. ESS) and cyclotrons (PSI) for neutron science or Accelerator Driven Systems. In the case of hadron colliders, magnet systems dominate electrical power consumption while RF systems only use a few percent. At CERN, where most accelerators are circular, only 6% of the annual consumption 1.1 TWh in 2017) are caused by the RF systems. For a 3 TeV CLIC machine, around 50% of the facility power consumption is due to the RF system. With an estimated yearly total of 2.74 TWh [4] and a European average price for non household consumption of 0.1e/kWh in 2017, this translates into 187 Me of electricity costs/year for the RF system alone. Therefore every percentage point in RF efficiency has a significant impact on the running cost. For Accelerator Driven Systems (ADS) the power to run the accelerator complex (including cooling, ventilation, cryogenics, offices, etc. must be small (5 10% compared to the power, which is produced in the ADS core. To achieve these values the accelerator complex must have a total efficiency between 0.2 eacc 0.4 [7] which presents a challenge for today's accelerators. If not achieved, the idea frank.gerigk@cern.ch Table 1: RF Power for Various Future Accelerators Project Ptotal [MW] PRF [MW] Pbeam [MW] PSI* 10 4 1.3 LEP2+ 120 42 19 FCC ee+ tbd 165 100 ESS* 35 15 5 CLIC 500* 272 109 9.8 ILC 500* 164 68 9.4 CLIC 3000* 582 289 28 total power is for the whole facility with experiments, +without injector of using ADS for energy production is unlikely to become economically viable [8] TODAY'S TECHNOLOGIES NEW DEVELOPMENTS In the following we will review today's state of the art of high power RF sources (see Table 2) and point to developments, which may change their efficiencies or power reach in the near future. As pulsed and Continuous Wave (CW) operation pose different challenges to the RF sources both modes of operation will be treated. Modulator Efficiency Most high power RF sources need a High voltage (HV) modulator, which transforms the voltage of the electrical grid to the HV and pulse pattern needed by the RF source. For CW operation in general but also for pulsed operation of gridded tubes (tetrodes, IOTs) the modulator is basically a HV power supply. Pulsed power for gridded tubes is obtained by simply pulsing the grid using the input RF signal, whereas HV pulses for the operation of non gridded tubes (klystrons, magnetrons) have to be formed by the modulator itself. Various topologies have been developed in recent years, which are adapted to different pulse lengths and voltage needs [9] i) Traditional pulse transformer based modulators with efficiencies between 85% and 90% and rise times in the range of a few hundred microseconds. ii) High frequency transformer based modulators such as the resonant polyphase design employed at SNS [10] or the stacked multi level (SML) design developed and used by the European Spallation Source (ESS) with efficiencies up to 92% and with a very short rise time in the range of 100 ms. iii) Transformerless modulators such as the Marx generator [11] which have the potential for even higher efficiency, or direct switch designs. 9th International Particle Accelerator Conference IPAC2018, Vancouver, BC, Canada JACoW Publishing ISBN: 978 3 95450 184 7 doi:10.18429/JACoW IPAC2018 MOYGB1
author_names: Frank Gerigk
title: Status and Future Strategy for Advanced High Power Microwave Sources for Accelerators
year: 2018

abstract: RF simulations of two different methods of implementing power combining of eight, 1kW, 500 MHz solid state power amplifiers are presented. The first using eight circulator plus isolation loads feeding an 8 way impedance combiner. The second method utilizes an eight input Gysel type power combiner. The overall electrical performance, response to the loss of one or two amplifiers, and estimated costs of each method is also compared to illustrate the issues involved. Introduction The steady progress of Solid State Power Amplifier technology has made the prospect of replacing high power tube transmitters more realizable in the 100 2000 MHz frequency bands. The method used to combine the outputs of those SSPA's has become more important since there is a push to achieve high efficiency and electrical performance while still protecting the output stages of the amplifiers. Ferrite circulators, feeding n port impedance power splitter/combiners, are the tried and true method of protection from high power reflections. However, at these output levels, there are significant impacts to both the insertion loss and power dissipation of using circulators. Using a corporate feed/combiner network of isolated 4 port devices is also an established method to combine these power levels. However, with the number SSPA's required to generate the 10 60kW final output range to compete with tube transmitters, this scheme becomes cumbersome and space inefficient. In addition, the isolation load power requirements get more demanding, and expensive, as the signal gets closer to the final output. The SSPA output power also makes the N port Wilkinson type combiners not usable due to the roughly 100W power dissipation limit on the isolation resistors. The N port Gysel type combiner offers an attractive alternative for combining multiple high power inputs. The configuration can be implemented in a space saving configuration, often in a 19" rack mountable package, and uses isolation loads that are on the same order as the power output of the SSPA. By using two levels of Gysel combiners, a system can easily be created to combine 50 to 80 SSPA outputs. The electrical performance can also be optimized to be very wideband (1 octave or more) for those applications that require that feature with high power. This paper compares two methods to specifically implement combining eight 1kW SSPA's at 500 MHz to illustrate some of the issues involved. The first uses eight, 1kW circulators (Type N coax inputs) feeding an 8way, coax impedance power splitter/combiner with a 3 1/8 output. The second method demonstrates an 8 input Gysel combiner also with Type N inputs and a 3 1/8 coax output. RF simulations establish the overall electrical performance of each method and the response of the two systems to a few possible scenarios of one or two of the SSPA's going dead or operating at reduced power. A comparison of the final implemented monetary cost of each system is also presented. Circulator with 8 way Splitter Method The analysis tool used to perform the RF simulations in this paper was Keysight's Genesys. The microwave circuit diagram is shown in Figure 1. You see from the diagram that the 8 way impedance power splitter is the star network of transmission lines (TL1 TL9) on the right side with the output being the 50 ohm port #10. The output quarter wave section is a 30.4 ohm section in 3 1/8 coax. The eight 86.6 ohm quarter wave sections feeding the output can be realized in any smaller sized coax that can handle the 1kW input power. This arrangement of quarter wave transformers gives a reasonably broad band splitter/combiner network by balancing the transition impedance ratio between any input and the output to the effective star point impedance. As a point of reference, if this network were viewed as and power splitter with port 10 as the input, then this port would have return loss of better than 20 dB from 465 to 535 MHz (14% BW) The center 8 symbols are the circulators with the load ports shown just below them starting with port 2 thru 9. For analysis purposes, these are ideal circulators with no insertion loss and infinite isolation. The remainder of the circuit symbols on the left hand side are ideal attenuators and phase shifters which are fed, in parallel by the ideal amplifier from port 1 with the gain set to 9.03 dB (gain of 8) This arrangement allows one to simulate 8 SSPA's feeding each circulator with 1kW of power. The attenuator and phase shifter allows individual control of the SSPA's output to simulate the effective power distribution given certain conditions. For example, as shown in Figure 1, if Attn_1 is set to 40 dB, this simulates the power loss of one of the eight SSPA's. The simulation then shows where the remaining power of the seven SSPA's is distributed. Figure 1. Schematic of 8 way Impedance Splitter/Combiner with circulators and feeding network Many different scenarios can thus be anticipated with this arrangement simply by applying different attenuations and phase shift distributions. For example, tolerances with respect to amplitude and phase balance of the SSPA's could be determined with respect to output power efficiency, or how to size the loads placed on the outputs of the circulators given the expected operating conditions or worst case scenarios. These simulations Impedance Splitter/Combiner with Circulators would be invaluable for the transmitter manufacturer to determine the operating conditions for the system of SSPA's. For the sake of comparison in this paper, 6 scenarios were simulated to illustrate the response of the combining networks. Table I gives the power output (in kW) at ports 2 thru 10 for the following conditions: 1. Nominal: All SSPA's in perfect amplitude and phase balance 2. 1 Amp 10o: One amplifier (port 2) with phase error of 10 degrees 3. 1 Amp 20o: One amplifier (port 2) with phase error of 20 degrees 4. 2 Amp 20o: Two amplifiers (ports 2 3) with phase error of 20 degrees 5. 1 Amp Dn: One amplifier (port 2) completely off, no output 6. 2 Amp Dn: Two amplifiers (ports 2 3) completely off, no output Port Nominal 1 Amp 10o 1 Amp 20o 2 Amp 20o 1 Amp Dn 2 Amp Dn #10 Main Output 8.000 7.973 7.894 7.761 6.142 4.530 #2 Refl. Power Thru Circulator .001 .023 .092 .119 .740 .543 #3 Refl. Power Thru Circulator .001 .001 .002 .119 .017 .543 #4 Refl. Power Thru Circulator .001 .001 .002 .001 .017 .064 #5 Refl. Power Thru Circulator .001 .001 .002 .001 .017 .064 #6 Refl. Power Thru Circulator .001 .001 .002 .001 .017 .064 #7 Refl. Power Thru Circulator .001 .001 .002 .001 .017 .064 #8 Refl. Power Thru Circulator .001 .001 .002 .001 .017 .064 #9 Refl. Power Thru Circulator .001 .001 .002 .001 .017 .064 Table I: Distribution of SSPA power (in kW) under various conditions for Circulator Splitter/Combiner Network One of the first observations to note is that phase imbalances of less than 10o do not cause much degradation in the output power in terms of percent power loss. The 10o phase imbalance of one SSPA causes less than 0.3% power reduction at the output. The majority of miss directed power comes back at the SSPA that is out of balance which is shunted by the circulator to its associated load (note that this will be a common phenomena of these combining networks) Even a 20o imbalance of one or two SSPAs cause a power reduction of 1.3% and 2.9% respectively. More simulations could be run to determine an RMS phase error that is required to achieve a certain efficiency goal, and thus, the requirements of the "drive" phase compensation network on the input of the SSPAs. The impact of having a SSPA lose power all together is much more dramatic, however. Note, the power from the seven remaining amplifiers will not all go towards the output. In fact, only 87.7% of the remaining 7kW will be seen at the output. The remainder is lost in the circulator loads, and most of that power is directed towards the SSPA that is down. Thus, if the intention is to still operate the system with one SSPA lost, then the designer must size the loads on the circulator to handle at least 75% of the SSPA output. The situation is even worse when two SSPAs are down: 75.5% of the 6kW power to the output, the rest to the circulator loads. Of course, the situation is different, if instead of an SSPA being total lost, one or two amplifiers run at reduced power. This situation was not presented but can be inferred from the data. For example, if one amplifier is running at 750W compared to the 1kW for the other seven, then 87.7% of that difference (250W) would go to the circulator load for that SSPA in the form of reflected power (i.e. 0.877*250W=219W) Thus, the total output would then be 7531W, or a 2.8% reduction of the total available power. With any of the above scenarios, the combiner system would still allow the SSPA system to operate at predictable reduced power with the circulators protecting all individual SSPAs from high power reflections. Note, this continued operation necessitates being able to dissipate the additional heat generated from the loads. One must also take note, that even with all the SSPAs operating at identical power and perfect phase balance there is still the practical insertion loss of the circulators which for these power levels are of the order of 0.2 to 0.5 dB which corresponds to 5 to 10% power lost which is the major drawback of this scheme of combining. 8 Input Gysel Combiner The Gysel circuit diagram is shown in Figure 2. The star network on the output side is essentially the same as an impedance splitter/combiner. The important difference is the additional star network to the left comprised of transmission lines TL3, TL7 13 and a set of intermediate quarter wave lines, TL2, TL6, TL14 19. It is this additional network of transmission lines that makes the input to the Gysel combiner matched independent of the condition of the other inputs. Much like how a 4 port passive device can have
author_names: P S Matthews
title: Comparison of Combining Schemes for 1 2kW Solid State Power Amplifiers in the 0.1 2 GHz Frequency Band
year: 2015

abstract: Power amplification at microwave frequency and above is one of strategic targets under investigation of vacuum microelectronic devices, because it has possibilities to generate more output power and higher frequency than solid state devices. Most of this work has focussed on a triode structure commonly based on a field emission array (FEA)l) FEA is sufficiently nonlinear in I V characteristics associated with Fowler Nordheim equation, which provides highly efficient operation, as expected from B and C class amplifiers. This is an additional advantage in EA based devices. FEA driven by RF power emits a pulsed electron beam with narrow width repeated at the RF signal. When the beam is sufficiently accelerated and coupled with a RF circuit, one can create a high power and high efficiency amplifier having a simple structure without the velocity modulation required in a multicavity klystron. Additionally, highly efficient harmonic amplification up to a higher order becomes possible by coupling with a harmonic RF circuit2) The paper describes the expected operation for the tubes. Figure 1 shows a schematic drawing of the microwave amplifier or frequency multiplier using the FEA. Figure 2 shows a phase distribution of emitted electron current density calculated for a RF FEA with tip density of 1.6x107A/cm2 by using the WKB approximation under a suitable tip shape. The emitted electrons are well bunched within a narrow range of RF phase, as shown in Fig. 2. Table 1 shows the peak efficiencies and output powers for the fundamental and higher harmonic operations. Here, the cathode diameter of lmm and an accelerating voltage of 30kV are assumed. Extraordinarily high efficiencies of 97% 90% 78% and 50% are expected at the fundamental, second, third and even fifth harmonics, respectively, and several tenth kW of output power are extracted from the simple and light weight tube. Operation frequency is commonly limited by the cut off frequency governed by transconductance and the parasitic capacitance of the FEA. However, there are many method to overcome the frequency limitation. For examples, since FEA is installed in a reentrant cavity at high frequency operation, a field emission may be extracted by RF field added on DC bias or basically no extraction electrode may be required for RF FEA operation. Furthermore, distributed interaction structure and Bragg resonator structure with or without gate electrode may be considered at millimeter waves and optical waves3)
author_names: Kuniyoshi Yokoo

Hidetaka Shimawaki

Shoichi Ono
title: Proposal Of A High Efficiency Microwave Power Source Using A Field Emission Array
venue: [Proceedings] IVMC '93 Sixth International Vacuum Microelectronics Conference
year: 1993

abstract: This paper describes the theory and development of a high power solid state amplifier technology for RF and microwave frequencies. The amplifier, which contains a large number of transistors, utilizes a pair of resonant cavities that efficiently function as input distributor and output combiner. Moreover, the modal structure and the high intrinsic Q of the cavities provide the required high transformer ratio for i/o matching and low insertion loss. The physical cavities can serve as heat sink, and also provide dc grounding and RF shielding. Experimental results show, among other desirable features, a 3dB bandwidth of 10% and a combining efficiency in the high 90% range. The structural design lends itself to ease of manufacture.
author_names: Bernard R-S Cheo

Douglas A Davids

Li-hsiang S Cheo

M K Ke
title: A Cavity Based High Power Solid State RF/Microwave Amplifier
venue: 2006 IEEE Sarnoff Symposium
year: 2006

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author_names: Bin-yi Chen
title: GSM/DCS Si Pin Shou Ji Gong Lu Fang Da Qi Mo Zu Dian Lu Te Xing Yan Jiu
year: 2013

abstract: The multiple beam, Ku band active phased array antennas currently being developed for communications satellites require broadband, reproducible, and power efficient solid state power amplifiers (SSPAs) Since these amplifiers will operate in a multiearricr environment, they must be designed to meet a stringent linearity specification while functioning at high efficiency. The design and development of a fully monolithic 1 W SSPA, and the production of SSPA modules for a 64 clement phased array antenna application, are described. A summary of phased array antenna systems considerations is included. Monolithic microwave integrated circuit (MMIC) design approaches and the thermal analysis of the SSPA module are also reviewed. The measured RF performance of the SSPA shows 22.5 dB small signal gain, output power of +30 dBm, and two tone third order intermodulation, C/I3, greater than 16 dB, The power added efficiency of the overall 1 W amplifier is greater than 25 percent, with a power chip efficiency greater than 28 percent. The measured RF performance of 80 SSPA modules is summarized in the form of histograms. The excellent performance uniformity obtained over all modules is the result of processinsensitive MMIC designs, chip selection procedures using on wafer DC and RF probe data, and a repeatable module assembly approach. Substantial portions of this paper appeared in the Proceedings of the AIAA 15th International Communications Satellite Systems Conference, San Diego, California, February /March 1994, A Collection of Technical Papers, Pt I. pp. 316 322. AIAA Paper No. 94 0952 CP.
author_names: Ramon A Khalona

G C Estee
title: Performance of M ary FSK modulation in a shadowed land mobile satellite communications channel
year: 2006

abstract: The VIL410 HeatwaveTM Inductive Output Tube (IOT) amplifier system has been developed to meet the requirements of superconducting RF accelerators. Two VIL410 systems were completed and delivered in April 2014. The VKL9130A1 IOT in the VIL410 provides up to 30 kW RF output power over a 5 MHz bandwidth centered at 1.30 GHz. It operates both CW and pulsed. The VIL410 amplifier has been designed to achieve very tight amplitude and phase control. The amplitude and phase ripple are specified to be less than 0.1% rms and better than 0.2deg rms, respectively. The stability of the output power is specified to be better than 0.2% over a 20 second period. In normal system operation, smooth control of the output power is accomplished via input from the low level RF system. The VIL410 uses CPI's VSL3616 Solid State Power Amplifier (SSPA) to drive the IOT. The VSL3616 is a 700 watt CW SSPA that operates at 250 watts CW in the VIL410. The VIL410 has an embedded processor that controls all internal functions of the amplifier system and interfaces directly to EPICS. The VIL410 can be operated locally using a LabViewTM PC Host program or remotely by EPICS. Figure 1: VIL410 IOT amplifier. RF REQUIREMENTS In order to operate into super conducting cavities, the VIL410 (Figure 1) must be protected by a four port waveguide circulator. The circulator protects the amplifier from high reflected RF power. The circulator is equipped with water loads that can absorb the full output power, allowing the amplifier to operate into an RF short of any phase without damage or fault shutdown. The circulator, manufactured by Ferrite Microwaves Technologies, LLC, was provided as part of the VIL410 system. The VIL410 meets the specifications listed in Table 1. Table 1: Specifications Parameter Specification Frequency 1300 MHz 3 dB Bandwidth 5 MHz Gain 72 dB Output Power 25 kW Input Power 1 mW Phase Variation 8deg Gain Variation 2 dB Amplitude Ripple 0.1 RMS Phase Ripple 0.2deg RMS Power Stability 0.2 in 20 sec Interval RF CHAIN The RF chain consists of an SSPA (VSL3616) driving an IOT (VKL9130A1) and then finally a four port WR650 differential phase shift circulator. The SSPA provides 250 W of drive power to the IOT with 62 dB gain typical input drive is 9 dBm. The IOT gain at 25 kW output power is 22 dB. The insertion loss of the circulator is 0.4 dB at full power. The overall system gain is 83.6 dB. VSL3616 SSPA The SSPA consists of a three stage driver amplifier assembly driving a two way in phase power splitter. The output from the splitter drives a pair of two stage power sections consisting of a 40 W stage driving a pair of 200 W unmatched power transistors operating in a balanced configuration. RF from the two pair of output devices is combined using a four way Gysel power combiner that has low insertion loss and high port to port isolation. The combined output provides up to 750W of RF power into a ferrite isolator that protects the SSPA from reflected power. The output of the isolator drives a dual directional coupler with a rear panel 7/16 DIN connector. The coupled outputs are used for forward and reverse power monitoring. The entire system including the DC power supplies is mounted to a water cooled plate to draw heat from the major assemblies. The system is contained in a 19 inch rack which is 4 units high and 26 inches deep. The unit is powered by 240 Vac and requires approximately 1900 W of power at full RF output. The VSL3616 SSPA is described in detail in [1] VKL9130A1 IOT Shown in Figure 2, the CPI VKL9130A1 selected for this transmitter achieves an output power of 30 kW CW ____________________________________________ #todd.treado@cpii.com Proceedings of IPAC2014, Dresden, Germany WEPME023 07 Accelerator Technology Main Systems T08 RF Power Sources ISBN 978 3 95450 132 8 2305 C op yr ig ht (c) 20 14 C C B Y3. 0 an d by th e re sp ec tiv e au th or s and can be operated CW or in pulsed mode. The IOT is a hybrid vacuum electron device incorporating a powergrid triode input with an inductive, klystron type output. Its beam is RF modulated by a gridded gun that biases the beam so that the amplifier can be adjusted for differing classes of operation. An IOT based amplifier has the advantage of possessing smaller foot print, lower capital cost, and enhanced power efficiencies when compared with 30 kW solid state or klystron based amplifiers. Figure 2: VKL9130A1 IOT. RF PERFORMANCE Gain and phase pushing as a function of output power are shown in Figure 3 for the VIL410. For output power between 5 and 25 kW, the gain and phase are wellbehaved. RF output power and efficiency are shown in Figure 4 as a function of input RF power. The efficiency of the VIL410 is 40% at 25 kW. The bandwidth of the VIL410 system is determined by the IOT amplifier and measured data is shown in Figure 5. It can be seen that the system meets a 2.5 MHz requirement. RMS amplitude and phase ripple were measured in a 100 mS window using 120 kHz bandwidth. The measured amplitude ripple was 0.1% RMS, and the phase ripple was 0.05deg RMS. The bridge configuration shown in Figure 6 was used to make these measurements. RMS ripple values, both phase and amplitude, were calculated from mixer I and Q waveforms that were digitized using a 12 bit sampling oscilloscope. Power stability was measured in a 20 second window. Because our cooling water regulator might cycle during the data acquisition, the stability numbers varied between measurements but were typically about 0.13% The measured phase sensitivity to coolant water temperature is 0.58deg/degC, and the power sensitivity is 0.07dB/degC. Figure 3: Gain and phase vs. output power. Figure 4: Output power and efficiency vs. input RF power. Figure 5: Gain vs. frequency. Figure 6: Measuring phase and amplitude ripple. WEPME023 Proceedings of IPAC2014, Dresden, Germany ISBN 978 3 95450 132 8 2306 C op yr ig ht (c) 20 14 C C B Y3. 0 an d by th e re sp ec tiv e au th or s 07 Accelerator Technology Main Systems T08 RF Power Sources Pulsed Operation Although the VIL410 system was designed for CW operation, it can be operated in pulsed mode by applying a pulsed RF input signal [2] In pulsed operation the peak power is still 25 kW and any pulse width greater than 100 ms is supported. Figure 7 shows a detected RF pulse 100 ms wide (top) Because the HV and grid power supplies are already on, the RF rise and fall time are only 120 ns, when measured from 10 to 90% (bottom) Figure 7: Pulse mode detected output RF. CABINET LAYOUT The VIL410 occupies three standard rack mount cabinets. The first cabinet houses two high voltage DC power supplies and a controller that synchronizes the outputs to achieve high regulation and low ripple. Also in this bay are the embedded industrial controller and fast fault control electronics. Additionally auxiliary power supplies and three phase AC line power distribution is accomplished in this bay. The middle cabinet is the HV bay. The floating grid deck power supply, stored energy capacitor, arc protection, and current sensing circuitry are mounted here. The third cabinet is the RF bay. The VSL3616 SSPA and the VKL9130A1 IOT are mounted in this bay, together with a dual directional waveguide coupler. The WR650 waveguide circulator is located outside the amplifier cabinet. See Figure 8 for component locations. Figure 8: VIL410 cabinet layout and circulator. AC POWER AND WATER COOLING The AC prime power requirement for the VIL410 is 380 Vac, 50/60 Hz, 130 kVA, Y (3 phase plus neutral and ground) At an RF output power of 26.3kW, the AC line current is about 100 A per leg. Figure 9 shows AC prime power as a function of RF output power. The required cooling water flow is 18 GPM. Figure 9: AC line power vs. output power. CONTRIBUTORS TO RF STABILITY The greatest technical challenge was meeting the RF stability requirements. For 30 kW RF output, Table 2 shows estimated IOT phase and amplitude sensitivities for RF drive, grid voltage, and cathode voltage. Some general observations: Grid voltage exerts the greatest influence on phase and amplitude stability. Meeting the amplitude stability specification guarantees meeting the phase stability specification. RF drive from the SSPA takes the bulk of the total amplitude and phase stability budget. Because the gain of the SSPA is much higher than that of the IOT, RF stability of the SSPA dominates in the system stability budget. Table 2: IOT RF Sensitivities at 30 kW Parameter Amplitude Phase RF Drive 158 Pout/Pin 0.06deg/W Grid Voltage 300 W/V 1.7deg/W Cathode Voltage 2.1 W/V 0.5deg/kV
author_names: Elkin Igor

Russ Army

Paul D Brown

Steve Locke

R Rizzo

Robert Snyder

G A Solomon

Mike Tracy

Todd A Treado
title: VIL410, CPI's 1.3 GHz, 25 kW CW IOT Amplifier System
year: 2014

abstract: 1 Review of semiconductor physics and devices. Energy bands. Statistical properties of electrons and holes. Carrier transport. Carrier recombination and generation. P N junctions. Schottky barriers. Reverse break down. Phonons. References. Further reading. 2 Transferred electron devices (TED) GUNN devices. Electron transfer and negative differential mobility. High field dipole domains in GUNN devices. Modes of operation of GUNN devices. Indium phosphide transferred electron devices/ millimeter wave operation of TED's. Example: Growth rate of a high field dipole Domain the "equal areas" rule. Stationary domain at the anode. Problems, Chapter 2. References. Further reading. 3 IMPATT (Impact Avalanche Transit Time) devices. Operation of IMPATT devices physical discussion. Small signal theory of IMPATT device impedance. Estimate of the power conversion efficiency of IMPATT devices a simple large signal model. Doping profiles for IMPATT diodes. An analytical large signal model of IMPATT devices. Non steady state large signal models for IMPATT devices. Problems, Chapter 3. References. Additional reading. 4 Tunneling devices. Tunnel diodes. Resonant tunneling devices. Problems, Chapter 4. References. Further reading. 5 Fundamental limitations on power output from solid state microwave devices. The thermal limit. The electronic limit. Measured data for rf power. Problems, Chapter 5. References. 6 Basic properties and circuit aspects of oscillators and amplifiers based on two terminal devices. A basic oscillator model. Injection locking of oscillators. Model for FM and AM noise in oscillators. Actual noise observed in two terminal solid state devices. Electronic tuning of solid state oscillators. Examples of actual circuits and impedance diagrams for GUNN and IMPATT oscillators. Negative resistance devices used as amplifiers. Problems, Chapter 6. References. Further reading. 7 Power combining. Chip level power combining. Circuit level power combining. Spatial (quasi optical) power combining. Problems, Chapter 7. References. Further reading. 8 Review of noise processes and noise concepts relevant to microwave semiconductor devices. Thermal noise noise figure and equivalent noise temperature. Shot noise. Diffusion noise. Flicker noise, or 1/F noise. References. Further reading. 9 Diode applications to microwave frequency conversion and control. Frequency conversion devices. Semi conductor diode detectors. Schottky barrier diodes. Semi conductor diode mixers: intrinsic conversion loss. Parasitic element effects in semiconductor mixers. Noise figure/noise temperature of mixer receivers. Other types of mixers. Noise temperature versus frequency for mixers. Varactor harmonic multipliers. PIN diodes and microwave control devices. Problems, Chapter 9. References. Further reading. 10 MESFET Devices. I V characteristics of MESFETs. Small signal equivalent circuit model. Ultra fast electrons, or how ballistic can an electron be. The Fukui noise model for MESFETs. The Pucel Haus State noise model. Noise in FET oscillators. Power frequency limitations in MESFETs. Overview. Problems, Chapter 10. References. Further reading. 11 HFETs Heterojunction Field Effect Transistors. Discussion of the I V characteristics of a HFET. Transconductance and cut off frequencies for HFETs. Indium based heterostructures for HFETs. Microwave equivalent circuit for HFETs. Noise modeling of HFETs comparison with MESFETs. Review of noise data for HFETs and MESFETs. HFET power amplifiers. HFET oscillators. Overview. Problems, Chapter 11. References. Further reading. 12 Bipolar microwave transistors. Basic relations for microwave BJTs. Equivalent circuit of the BJT frequency performance. Noise modeling of BJTs. BJT power amplifiers and oscillators. Heterojunction bipolar transistors (HBTs) Structure and I V characteristics of HBTs. Equivalent circuit and cut off frequencies of HBTs. HBTs with other material combinations than A1GaAs/GaAs. Noise properties of HBTs. HBT power amplifiers and oscillators. Overview. Problems, Chapter 12. References. Further reading. 13 Overview of conventional and novel devices. Hot electron transistors. Resonant tunneling transistors. Permeable base transistors. Review of the performance of microwave semiconductor devices 1990. Conclusion. References. Further reading.
author_names: K Sigfrid Yngvesson
title: Microwave semidonductor [i.e. semiconductor] devices
year: 1991