# Compact and Multiband MIMO Dielectric Resonator Antenna for Automotive LTE Communications

Authors: Tzu-Ling Chiu; Laure Huitema; Olivier Pajona; Thierry Monediere
Journal: International Journal of Antennas and Propagation
 (2018)

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## Abstract

A compact and multiband dielectric resonator antenna (DRA) designed for LTE automotive solutions is presented in this paper. The proposed MIMO system is located on the vehicle rooftop within a limited space of 120 mm × 70 mm × 65 mm. To cover all the LTE standard frequency bands used around the world, the antenna is matched around 790 MHz–860 MHz, 1700 MHz–2200 MHz, and 2500 MHz–2700 MHz frequency bands with a∣S11∣ lower than −6 dB while presenting a minimum total efficiency of 50% with a maximum realized gain better than 1 dB on all these frequency bands. The DRA is then mounted and measured on a real vehicle rooftop in order to see its performances in real operation conditions. Finally, to improve both the quality and reliability of the wireless link, two DRAs are placed within a small area to reconfigure their radiation patterns on each frequency band. Measured performances, which are in good agreement with the simulated results, are used to validate if the antenna design is suitable for LTE MIMO systems to be integrated on an automotive. The MIMO system is evaluated using the envelope correlation coefficient (ECC), and the obtained value for the proposed antenna is lower than 0.25.


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## Body

## 1. Introduction

The explosive growth of wireless communications and the increasing demand for high data rate throughput push the deployment of Long Term Evolution (LTE) communication systems. From the beginning, the LTE standardization effort focused on enhancing Universal Terrestrial Radio Access (UTRA) and optimizing the Third Generation Partnership Project’s (3GPP’s) radio access architecture. Its initial release was finalized in 2008, and now, the LTE standard works on four frequency bands all over the world [1].The Multiple Input–Multiple Output (MIMO) technique is one of the key technologies in the LTE system. Indeed, a MIMO system can help to overcome the damage effects of multipath. It is now a commonly used method to increase the downlink and uplink data rates without additional spectrum allocation.Several printed multiband MIMO antenna systems, based on the LTE standards for wireless devices [2, 3], have been published in the literature in the past few years. Among these papers, only a few of them are focusing on the lowest LTE band, which operates around 800 MHz. Indeed, at these frequencies, the antenna needs to be electrically small to be integrated within the wireless devices. Nevertheless, it is well known that, due to fundamental physics limits described in [4], small antennas are limited, generally in terms of radiation performances (efficiency) and bandwidth (quality factor). In addition, the placement of many antennas (two in our case) close together increases the mutual coupling between them and affects the diversity performances of MIMO systems [5, 6].In [7], a MIMO system made of four cylindrical dielectric resonator antenna (DRA) elements was developed for the 2.45 GHz and 5.8 GHz WLAN bands. In [8], authors showed that with proper excitations for different characteristic modes of planar antennas, which are orthogonally placed, good matching and low coupling and correlation can be achieved for working frequencies lower than 1 GHz. In [9], a study on how to decrease the coupling of two cylindrical DRAs is presented at 10 GHz. Other solutions to decrease the coupling of a MIMO system are based on the excitation of orthogonal modes either by using printed antennas [10] or dielectric resonator antennas [11–13]. Another solution is to orthogonally integrate the two MIMO antennas [8].There are several feed mechanisms for the DRA [14] such as probe near the DRA [15], microstrip line [16], aperture-coupled [17], and dielectric image guide coupling [18]. Moreover, DRAs can have different shapes such as cylindrical, rectangular, pyramidal, spherical, and hemispherical. The shape of the DRA, its feeding mechanism, and its wall boundary condition play an important role in its operating frequencies and impedance bandwidths.In [19, 20], we presented the simulation of a beam steering system for an automotive application. In this framework and to steer the beam, a phase shifter has been designed and integrated. Now, in this paper, a MIMO LTE system having a pattern diversity is developed. The antenna development is based on modal analyses in order to cover all the LTE bands while being miniature. Resonance frequencies of a rectangular-shaped DRA are studied. Some faces can be completely or partially coated with a metal in order to obtain the wanted working frequencies. The shape must also be easy to manufacture for commercial purposes. The DRA is designed to cover the LTE spectrum operated on four distinct frequency bands: LTE 800DD (band 20), LTE 1800+ (band 3), LTE 2100 (band 1), and LTE 2600 (band 7). In order to have a MIMO system for the LTE standard, the antenna also needs to have reconfigurable radiation patterns in these four frequency bands.Based on commercial LTE wireless module specifications, a∣S11∣ lower than –6 dB with a minimum total efficiency of 50% on the four frequency bands is required. The MIMO antenna system performances are evaluated with the envelope correlation coefficient (ECC). A value under 0.5 is required.The proposed MIMO system is designed to be integrated on the rooftop of a vehicle within a limited space of 120 mm × 70 mm × 65 mm corresponding toλ0/3×λ0/5×λ0/6 at 800 MHz as presented in Figure 1.Figure 1
Dedicated space of automotive antenna.The roof of the car is considered as a large ground plane. For the ease of measurement and manufacture, we initially used a 200 mm × 200 mm-FR4 substrate entirely coated by copper. Finally, the prototype was mounted and measured inside an anechoic chamber on a car rooftop. The corresponding performances are presented.


## 2. Antenna Design and Configuration

### 2.1. DRA Development

To integrate two antennas within the allocated space, each radiating element needs to be miniaturized. In this framework, we set the dimensions of one dielectric resonator (DR) at 22 mm × 13 mm × 50 mm to be vertically placed on a finite FR4 substrate (Figure2). Antenna’s characteristics (dimensions, quality factor, bandwidth, gain, and efficiency) depend on both the relative permittivity and the loss tangent of the dielectric material. The selected dielectric resonator relative permittivity is chosen close to 10 in order to obtain the widest impedance bandwidth, while having the smallest possible dimensions. Therefore, the chosen DR material is the Rogers TMM10 with a relative permittivity of 9.2 and a loss tangent of 0.0022.Figure 2
Maximum dimensions of the antenna to be integrated within the dedicated space presented in Figure1.For the modal analysis, the FR4 substrate is considered to have the same lateral dimensions as the dielectric resonator (see Figure2) with a copper metallic ground on the bottom side. Its relative permittivity is 4.1 with a loss tangent of 0.012 at 1 GHz. Within these dimensions and considering a dielectric resonator placed on a FR4 substrate coated by copper, the first resonant mode is the TE101 at 2 GHz (Figure 3).Figure 3
(a) E-fields on they=11 mm plane of an isolated DRA placed on a FR4 substrate coated by copper. (b) Integrating a vacuum cavity surrounded by PMC conditions during eigenmode analyses.
(a)
(b)It should be noticed that this resonance frequency has been determined considering evanescent modes at the interface between the dielectric resonator and the air, i.e., without considering a perfect magnetic condition (PMC). The metallic conditions are obtained using copper layers. To proceed, a vacuum cavity surrounded by PMC conditions is integrated around the DRA during the eigenmode analyses (Figure3(b)). The thickness of the cavity is chosen such that we obtain a convergence of the resonance frequency of the mode with eigenmode analyses. These eigenmode analyses are then validated by the electromagnetic simulation by integrating an excitation port and checking both the resonance frequencies and the 3D shapes of each mode.This mode is presenting a symmetry in thex=6.5 mm plane. Therefore, the size of the DR can be halved by placing a metal plate in this plane while keeping the same resonant mode. It also means that if the global dimensions of the DRA are kept while integrating a copper coating at x=13 mm, the resonance frequency of the mode will decrease at 1 GHz. This second solution has been chosen in order to reach the lowest LTE band. The modal analysis is presented in Figure 4(a). From this modal analysis, when considering the metallic plate, the E-field is mostly concentrated on its upper side. Therefore, this metallic plate can be cut without excessively disturbing the mode. Figure 4(b) shows that the field inside the DR with the metallic plate halved remains mostly the same.Figure 4
E-fields on they=11 mm plane of a DRA placed on a FR4 substrate coated by copper, with a metallic plate at xmax (a) and with a partial metallic plate at xmax (b).An excitation port is then integrated in order to compute the antenna input impedance and make the analogy with the previous eigenmode resolution, which did not integrate an excitation. In this framework, input impedances for these three previous cases have been plotted in Figure5. As expected, the fundamental mode of the DRA without metallic condition (Figure 5) has its resonance frequency twice higher than the DRA with the metallic condition. This resonance frequency is little when the lateral face is partially coated by metal since it is resonating at 1.1 GHz for the cut metal plate condition (Figures 4(b) and 5). Starting from the metal plate condition, we can change the metallic part shapes in order to create hybrid modes between the two first modes at 1.1 GHz and 2.85 GHz, respectively (Figure 5).Figure 5
Input impedances for different DRA boundary conditions.Some parametric studies on the metallic plate showed that we can create two hybrid modes without disturbing the first resonating mode. On Figure6, the proposed antenna structure is detailed and shows the surface where metallic areas are integrated. Table 1 details the DRA dimensions.Figure 6
Geometry of the proposed DRA unit (a) in 3D view, (b) side view for SMA connector location, and (c) top view for matching circuitry connection of DRA.
(a)
(b)
(c)Table 1
Design parameters of the proposed antenna.
Parameter
a
b
c
d1
d2
Value (mm)
13
22
50
109
20
Parameter
w1
w2
w3
w4
w5
w6
Value (mm)
4.5
6.3
13.5
3.4
7
4
Parameter
h1
h2
h3
h4
h5
h6
h7
Value (mm)
9.7
2
24.3
14
20.5
8
14.5The input impedance of the optimized design with the corresponding modal analysis is presented in Figure7.Figure 7
Input impedance with the modal analyses (E-fields) for the final design.Since this antenna is developed to be integrated on an automobile rooftop, it is studied on a 200 mm × 200 mm-FR4 substrate entirely coated by copper. Figure7 is the DRA response on the proposed 200 mm × 200 mm ground plane. For higher frequencies, both eigenmode analyses of the DR and the input impedance presented in Figure 7 show that the parasitic strip on the side wall creates hybrid modes. From modal analyses presented in [21–23] and regarding the 3D fields from the eigenmode study, the excited hybrid modes (corresponding to the second and third modes) at 1.8 GHz and 2.42 GHz are TE3/201 and TE3/211, respectively.The resonance frequency of the fundamental mode is only slightly shifted compared with the metal plate cut condition since it is equal to 1.18 GHz. The integration of a large ground plane means that the global dimensions of the radiating element are increasing, which involves the decrease of theQ factor. Indeed, from the input impedance, we can determine the antenna quality factor by using equation (1) [24] where ω0 is resonant angular frequency, R0 is the resistance, and Z0 is the input impedance:
(1)Qω0=2R0ω0ω0Z0ω0.TheQ factor is reaching a maximum value of 30 at 1.04 GHz for the antenna on the 200 mm × 200 mm ground plane while it equals 100 for the DRA alone (without a large ground plane). It is lower than 5 for the other modes of the antenna mounted on the ground plane. A high Q factor implies that it is difficult to match the antenna on a wide bandwidth.The interest of exciting hybrid modes in this design is that the real part of the input impedance remains around 50Ω while having an imaginary part around 0 Ω between 1.4 GHz and 2.6 GHz. A wider impedance bandwidth is therefore obtained compared to the other cases, as presented by the ∣S11∣ parameters on Figure 8.Figure 8
∣S11∣ parameters for the four studied cases.It should be reminded that the antenna needs to work on the lowest LTE band between 790 MHz and 860 MHz. Therefore, a T-type matching circuitry is placed on the FR4 substrate just before the DRA to match the antenna on the serial resonance of the first mode, i.e., at 800 MHz. This matching circuit is presented in Figure9(a). Measured touchstone files with a homemade TRL kit for inductor and capacitor from Murata LQG and GRM series are used in the electromagnetic simulation.Figure 9
(a) Matching circuitry of DRA. (b) Photograph of the antenna.
(a)
(b)


### 2.2. DRA Performance

To validate the design, a prototype of the DRA was built as shown in Figure9(b). Its performances were measured in an anechoic chamber.Measurements were performed from 0.7 GHz to 2.7 GHz, where the four LTE bands are operating. Simulated and measured results are in a good agreement, and they exhibit a broadband and multiband DRA since it covers 790 MHz–860 MHz, 1575 MHz–2200 MHz, and 2500 MHz–2700 MHz frequency bands with a∣S11∣ lower than −6 dB as shown in Figure 10.Figure 10
Measured and simulated∣S11∣ parameters for the proposed DRA.3D radiation patterns have been measured inside an anechoic chamber; they are compared with the simulated results in Figure11 on three different planes, i.e., the x0y plane (θ=90o), x0z plane (φ=0o), and y0z plane (φ=90o), at the center of the four LTE frequency bands. The cross-polarized field is high, and the antenna is not exhibiting a linear polarization due to the DRA location, which is close to the edge of the ground plane. However, a pure polarization in this design is not needed since LTE specifications depend on the total radiated field Etot=Eθ2+Eφ2, including both the co- and the cross-polarizations.Figure 11
Simulated and measured 2D radiation patterns for the proposed DRA.For LTE antennas in automobile applications, it is an advantage if the radiation pattern for all azimuth angles shows an omnidirectional shape. Figure11 demonstrates that the proposed antenna can have the almost omnidirectional radiation pattern in the x0y plane for four LTE bands.From the 3D radiation patterns, we can deduce both the maximum realized gain and the total efficiency according to the frequency as presented in Figure12. Measurement results are in good agreement with the simulated ones and exhibit that the antenna efficiency is greater than 50% for the first LTE band around 800 MHz and better than 75% for the other three bands.Figure 12
Measured and simulated maximum realized gains and total efficiencies for the proposed DRA.A radome, with the geometry described in Figure1, is finally integrated and enclosed the antenna. The used material is a painting plastic. Figure 13 compares the measured ∣S11∣ parameter and efficiency according to frequency with and without this radome. As expected, the antenna performances are more disturbed on the lowest band since the radome is closer in terms of λ0 at lower frequencies. Indeed, on the first LTE band, the working frequency is a little bit shifted to the lower frequencies, i.e., there is a 20 MHz frequency shift around 800 MHz. The antenna efficiency is a little bit lower than for the ideal case but remains higher than 50%.Figure 13
Radome effect on the∣S11∣ parameter (a) and on the efficiency (b).
(a)
(b)The DRA is then mounted on a real automobile (Figure14) and measured to determine the antenna efficiency and its radiation patterns in an anechoic chamber. Figure 15 shows the 2D simulated gain patterns compared to the measurement on the vehicle rooftop. The main beam direction for both low and high bands occurs at the elevation angle of θ=60° counting from the zenith. Figure 15 is therefore presenting the comparison of simulated and measured gain patterns at θ=60°.Figure 14
Photo of the measurement for DRA implement on the vehicle rooftop.Figure 15
2D radiation patterns of the DRA on the vehicle rooftop.The maximum realized gain for the required bands is around 4 dB (Figure16). Compared to the upper hemisphere maximum realized gain for the DRA on the 200 mm × 200 mm ground plane, the realized gain is a little greater since the ground plane can be considered as infinite in this case.Figure 16
Comparison of maximum realized gains according to the frequency.Therefore, the proposed DRA structure has been validated in measurement with performances fitting with the LTE wireless module specifications.


### 2.3. Performances versus the State-of-the-Art

In order to compare the performances of our fabricated dielectric resonator antenna, Table2 provides a comparison of our results with different papers [22–25-32] presenting antennas dedicated to LTE bands (at least for the lowest band starting at 790 MHz). Their dimensions and operating bands with their corresponding maximum total efficiencies and maximum realized gains are compared. Some of them are dedicated for an integration within a vehicle [26–32] and also for a MIMO application [22, 25, 26, 31, 32].Table 2
Comparison with other studies.
Reference
Electrical size
Matching bands (MHz)
Maximum total efficiency (dB)
Maximum realized gain (dBi)
[22]
λ0/15×λ0/38×λ0/94
790–862
−4
−1.5
2300–2500
−0.9
5
3200–3600
−0.9
6.5
[25]
λ0/6×λ0/5×λ0/9
828–880
−0.56
1.6
1600–2200
−0.23
4.3
[26]
λ0/7×λ0/7×λ0/12
828–880
N/A
1
1710–2170
1
2490–2690
1
[27]
λ0/5×λ0/5×λ0/12
780–2350
N/A
4
6
[28]
λ0/5×λ0/17×λ0/325
698–960
N/A
4
1710–2170
5.5
2490–2690
2.5
[29]
λ0/9×λ0/11×λ0/8
698–960
−1.3
5
1470–2690
−0.7
6
[30]
λ0/8×λ0/8×λ0/8
698–960
N/A
4.5
1470–2690
4.5
5850–5925
4.5
[31]
λ0/6×λ0/13×λ0/13
700–960
N/A
4
1700–2170
5.2
2500–2700
7
3400–3800
8.1
[32]
λ0/9×λ0/13
698–960
N/A
4
1710–2690
4
Proposed antenna
λ0/8×λ0/17×λ0/29
790–894
−0.97
2
1710–2170
−0.46
4
2490–2690
−0.97
4As observed, our reported antenna is a good trade-off between its dimensions and radiated performances. Moreover, it covers all the four distinct frequency bands: LTE 800DD (band 20), LTE 1800+ (band 3), LTE 2100 (band 1), and LTE 2600 (band 7). As the main objective of this study is to enhance the quality of the wireless link by employing the radiation pattern diversity technique, its low sizes and good efficiency imply that we can integrate two of the previously presented antennas within the dedicated volume defined in Figure1.


## 2.1. DRA Development

To integrate two antennas within the allocated space, each radiating element needs to be miniaturized. In this framework, we set the dimensions of one dielectric resonator (DR) at 22 mm × 13 mm × 50 mm to be vertically placed on a finite FR4 substrate (Figure2). Antenna’s characteristics (dimensions, quality factor, bandwidth, gain, and efficiency) depend on both the relative permittivity and the loss tangent of the dielectric material. The selected dielectric resonator relative permittivity is chosen close to 10 in order to obtain the widest impedance bandwidth, while having the smallest possible dimensions. Therefore, the chosen DR material is the Rogers TMM10 with a relative permittivity of 9.2 and a loss tangent of 0.0022.Figure 2
Maximum dimensions of the antenna to be integrated within the dedicated space presented in Figure1.For the modal analysis, the FR4 substrate is considered to have the same lateral dimensions as the dielectric resonator (see Figure2) with a copper metallic ground on the bottom side. Its relative permittivity is 4.1 with a loss tangent of 0.012 at 1 GHz. Within these dimensions and considering a dielectric resonator placed on a FR4 substrate coated by copper, the first resonant mode is the TE101 at 2 GHz (Figure 3).Figure 3
(a) E-fields on they=11 mm plane of an isolated DRA placed on a FR4 substrate coated by copper. (b) Integrating a vacuum cavity surrounded by PMC conditions during eigenmode analyses.
(a)
(b)It should be noticed that this resonance frequency has been determined considering evanescent modes at the interface between the dielectric resonator and the air, i.e., without considering a perfect magnetic condition (PMC). The metallic conditions are obtained using copper layers. To proceed, a vacuum cavity surrounded by PMC conditions is integrated around the DRA during the eigenmode analyses (Figure3(b)). The thickness of the cavity is chosen such that we obtain a convergence of the resonance frequency of the mode with eigenmode analyses. These eigenmode analyses are then validated by the electromagnetic simulation by integrating an excitation port and checking both the resonance frequencies and the 3D shapes of each mode.This mode is presenting a symmetry in thex=6.5 mm plane. Therefore, the size of the DR can be halved by placing a metal plate in this plane while keeping the same resonant mode. It also means that if the global dimensions of the DRA are kept while integrating a copper coating at x=13 mm, the resonance frequency of the mode will decrease at 1 GHz. This second solution has been chosen in order to reach the lowest LTE band. The modal analysis is presented in Figure 4(a). From this modal analysis, when considering the metallic plate, the E-field is mostly concentrated on its upper side. Therefore, this metallic plate can be cut without excessively disturbing the mode. Figure 4(b) shows that the field inside the DR with the metallic plate halved remains mostly the same.Figure 4
E-fields on they=11 mm plane of a DRA placed on a FR4 substrate coated by copper, with a metallic plate at xmax (a) and with a partial metallic plate at xmax (b).An excitation port is then integrated in order to compute the antenna input impedance and make the analogy with the previous eigenmode resolution, which did not integrate an excitation. In this framework, input impedances for these three previous cases have been plotted in Figure5. As expected, the fundamental mode of the DRA without metallic condition (Figure 5) has its resonance frequency twice higher than the DRA with the metallic condition. This resonance frequency is little when the lateral face is partially coated by metal since it is resonating at 1.1 GHz for the cut metal plate condition (Figures 4(b) and 5). Starting from the metal plate condition, we can change the metallic part shapes in order to create hybrid modes between the two first modes at 1.1 GHz and 2.85 GHz, respectively (Figure 5).Figure 5
Input impedances for different DRA boundary conditions.Some parametric studies on the metallic plate showed that we can create two hybrid modes without disturbing the first resonating mode. On Figure6, the proposed antenna structure is detailed and shows the surface where metallic areas are integrated. Table 1 details the DRA dimensions.Figure 6
Geometry of the proposed DRA unit (a) in 3D view, (b) side view for SMA connector location, and (c) top view for matching circuitry connection of DRA.
(a)
(b)
(c)Table 1
Design parameters of the proposed antenna.
Parameter
a
b
c
d1
d2
Value (mm)
13
22
50
109
20
Parameter
w1
w2
w3
w4
w5
w6
Value (mm)
4.5
6.3
13.5
3.4
7
4
Parameter
h1
h2
h3
h4
h5
h6
h7
Value (mm)
9.7
2
24.3
14
20.5
8
14.5The input impedance of the optimized design with the corresponding modal analysis is presented in Figure7.Figure 7
Input impedance with the modal analyses (E-fields) for the final design.Since this antenna is developed to be integrated on an automobile rooftop, it is studied on a 200 mm × 200 mm-FR4 substrate entirely coated by copper. Figure7 is the DRA response on the proposed 200 mm × 200 mm ground plane. For higher frequencies, both eigenmode analyses of the DR and the input impedance presented in Figure 7 show that the parasitic strip on the side wall creates hybrid modes. From modal analyses presented in [21–23] and regarding the 3D fields from the eigenmode study, the excited hybrid modes (corresponding to the second and third modes) at 1.8 GHz and 2.42 GHz are TE3/201 and TE3/211, respectively.The resonance frequency of the fundamental mode is only slightly shifted compared with the metal plate cut condition since it is equal to 1.18 GHz. The integration of a large ground plane means that the global dimensions of the radiating element are increasing, which involves the decrease of theQ factor. Indeed, from the input impedance, we can determine the antenna quality factor by using equation (1) [24] where ω0 is resonant angular frequency, R0 is the resistance, and Z0 is the input impedance:
(1)Qω0=2R0ω0ω0Z0ω0.TheQ factor is reaching a maximum value of 30 at 1.04 GHz for the antenna on the 200 mm × 200 mm ground plane while it equals 100 for the DRA alone (without a large ground plane). It is lower than 5 for the other modes of the antenna mounted on the ground plane. A high Q factor implies that it is difficult to match the antenna on a wide bandwidth.The interest of exciting hybrid modes in this design is that the real part of the input impedance remains around 50Ω while having an imaginary part around 0 Ω between 1.4 GHz and 2.6 GHz. A wider impedance bandwidth is therefore obtained compared to the other cases, as presented by the ∣S11∣ parameters on Figure 8.Figure 8
∣S11∣ parameters for the four studied cases.It should be reminded that the antenna needs to work on the lowest LTE band between 790 MHz and 860 MHz. Therefore, a T-type matching circuitry is placed on the FR4 substrate just before the DRA to match the antenna on the serial resonance of the first mode, i.e., at 800 MHz. This matching circuit is presented in Figure9(a). Measured touchstone files with a homemade TRL kit for inductor and capacitor from Murata LQG and GRM series are used in the electromagnetic simulation.Figure 9
(a) Matching circuitry of DRA. (b) Photograph of the antenna.
(a)
(b)


## 2.2. DRA Performance

To validate the design, a prototype of the DRA was built as shown in Figure9(b). Its performances were measured in an anechoic chamber.Measurements were performed from 0.7 GHz to 2.7 GHz, where the four LTE bands are operating. Simulated and measured results are in a good agreement, and they exhibit a broadband and multiband DRA since it covers 790 MHz–860 MHz, 1575 MHz–2200 MHz, and 2500 MHz–2700 MHz frequency bands with a∣S11∣ lower than −6 dB as shown in Figure 10.Figure 10
Measured and simulated∣S11∣ parameters for the proposed DRA.3D radiation patterns have been measured inside an anechoic chamber; they are compared with the simulated results in Figure11 on three different planes, i.e., the x0y plane (θ=90o), x0z plane (φ=0o), and y0z plane (φ=90o), at the center of the four LTE frequency bands. The cross-polarized field is high, and the antenna is not exhibiting a linear polarization due to the DRA location, which is close to the edge of the ground plane. However, a pure polarization in this design is not needed since LTE specifications depend on the total radiated field Etot=Eθ2+Eφ2, including both the co- and the cross-polarizations.Figure 11
Simulated and measured 2D radiation patterns for the proposed DRA.For LTE antennas in automobile applications, it is an advantage if the radiation pattern for all azimuth angles shows an omnidirectional shape. Figure11 demonstrates that the proposed antenna can have the almost omnidirectional radiation pattern in the x0y plane for four LTE bands.From the 3D radiation patterns, we can deduce both the maximum realized gain and the total efficiency according to the frequency as presented in Figure12. Measurement results are in good agreement with the simulated ones and exhibit that the antenna efficiency is greater than 50% for the first LTE band around 800 MHz and better than 75% for the other three bands.Figure 12
Measured and simulated maximum realized gains and total efficiencies for the proposed DRA.A radome, with the geometry described in Figure1, is finally integrated and enclosed the antenna. The used material is a painting plastic. Figure 13 compares the measured ∣S11∣ parameter and efficiency according to frequency with and without this radome. As expected, the antenna performances are more disturbed on the lowest band since the radome is closer in terms of λ0 at lower frequencies. Indeed, on the first LTE band, the working frequency is a little bit shifted to the lower frequencies, i.e., there is a 20 MHz frequency shift around 800 MHz. The antenna efficiency is a little bit lower than for the ideal case but remains higher than 50%.Figure 13
Radome effect on the∣S11∣ parameter (a) and on the efficiency (b).
(a)
(b)The DRA is then mounted on a real automobile (Figure14) and measured to determine the antenna efficiency and its radiation patterns in an anechoic chamber. Figure 15 shows the 2D simulated gain patterns compared to the measurement on the vehicle rooftop. The main beam direction for both low and high bands occurs at the elevation angle of θ=60° counting from the zenith. Figure 15 is therefore presenting the comparison of simulated and measured gain patterns at θ=60°.Figure 14
Photo of the measurement for DRA implement on the vehicle rooftop.Figure 15
2D radiation patterns of the DRA on the vehicle rooftop.The maximum realized gain for the required bands is around 4 dB (Figure16). Compared to the upper hemisphere maximum realized gain for the DRA on the 200 mm × 200 mm ground plane, the realized gain is a little greater since the ground plane can be considered as infinite in this case.Figure 16
Comparison of maximum realized gains according to the frequency.Therefore, the proposed DRA structure has been validated in measurement with performances fitting with the LTE wireless module specifications.


## 2.3. Performances versus the State-of-the-Art

In order to compare the performances of our fabricated dielectric resonator antenna, Table2 provides a comparison of our results with different papers [22–25-32] presenting antennas dedicated to LTE bands (at least for the lowest band starting at 790 MHz). Their dimensions and operating bands with their corresponding maximum total efficiencies and maximum realized gains are compared. Some of them are dedicated for an integration within a vehicle [26–32] and also for a MIMO application [22, 25, 26, 31, 32].Table 2
Comparison with other studies.
Reference
Electrical size
Matching bands (MHz)
Maximum total efficiency (dB)
Maximum realized gain (dBi)
[22]
λ0/15×λ0/38×λ0/94
790–862
−4
−1.5
2300–2500
−0.9
5
3200–3600
−0.9
6.5
[25]
λ0/6×λ0/5×λ0/9
828–880
−0.56
1.6
1600–2200
−0.23
4.3
[26]
λ0/7×λ0/7×λ0/12
828–880
N/A
1
1710–2170
1
2490–2690
1
[27]
λ0/5×λ0/5×λ0/12
780–2350
N/A
4
6
[28]
λ0/5×λ0/17×λ0/325
698–960
N/A
4
1710–2170
5.5
2490–2690
2.5
[29]
λ0/9×λ0/11×λ0/8
698–960
−1.3
5
1470–2690
−0.7
6
[30]
λ0/8×λ0/8×λ0/8
698–960
N/A
4.5
1470–2690
4.5
5850–5925
4.5
[31]
λ0/6×λ0/13×λ0/13
700–960
N/A
4
1700–2170
5.2
2500–2700
7
3400–3800
8.1
[32]
λ0/9×λ0/13
698–960
N/A
4
1710–2690
4
Proposed antenna
λ0/8×λ0/17×λ0/29
790–894
−0.97
2
1710–2170
−0.46
4
2490–2690
−0.97
4As observed, our reported antenna is a good trade-off between its dimensions and radiated performances. Moreover, it covers all the four distinct frequency bands: LTE 800DD (band 20), LTE 1800+ (band 3), LTE 2100 (band 1), and LTE 2600 (band 7). As the main objective of this study is to enhance the quality of the wireless link by employing the radiation pattern diversity technique, its low sizes and good efficiency imply that we can integrate two of the previously presented antennas within the dedicated volume defined in Figure1.


## 3. MIMO System Structure and Performances

### 3.1. Structure of the MIMO Antenna System

The proposed MIMO system integrates two DRAs placed face-to-face in such a way that they can be embedded in the automotive dedicated space. Therefore, the distance between the DRAs is 18 mm along thex-axis and 40 mm for the y-axis. Each DRA is independently fed with a 50 Ω coaxial cable, and the system shape and element locations are symmetric as presented in Figure 17.Figure 17
MIMO antenna system in CST and prototype.


### 3.2. Performance of the MIMO Antenna System

S-parameters are presented in Figures 18(a) and 18(b). Since DRAs are similar and the structure is symmetric, the ∣S11∣ and ∣S22∣ parameters are almost similar, and they are well matched on the four LTE frequency bands. The system reciprocity implies that the ∣S21∣ parameter is equal to the ∣S12∣ parameter. Since the distance between the two radiating elements equals λ0/8.5 at 800 MHz, the coupling is reaching its maximum of −6 dB on the first working band. This coupling will therefore result in a lower realized gain and thus a lower efficiency compared to the antenna studied alone.Figure 18
MIMO system: (a) measured and simulated∣S11∣ and ∣S22∣ parameters and (b) measured and simulated ∣S21∣ parameters.
(a)
(b)Figure19 plots the total efficiency and maximum realized gain according to the frequency. As expected, it demonstrates that the antenna efficiency is lower for frequency bands where the coupling is higher than 10 dB. However, this efficiency remains more than acceptable since it is higher than 50% around 800 MHz, higher than 70% between 1.7 GHz and 2.2 GHz, and better than 60% for the highest LTE frequency band.Figure 19
Measured and simulated maximum realized gains and total efficiencies of the MIMO system.Figure20 shows the 3D measured radiation patterns for the main and the MIMO antennas at 820 MHz, 1575 MHz, 1800 MHz, 2150 MHz, and 2600 MHz. It can be observed that this device offers the possibility to perform a pattern diversity since the radiation pattern is depending on the excited DRA. Figure 21 shows the 2D radiation patterns in the x0y plane (θ=90°). There is a good agreement between the measurement and the simulation.Figure 20
Radiation patterns of the MIMO system in 3D measurement.Figure 21
2D radiation patterns of each DRA elements in the MIMO system.However, these results are not sufficient to conclude that the radiation pattern is reconfigured. An important parameter to evaluate the diversity performances of a system is the envelope correlation coefficient (ECC). Indeed, it evaluates the correlation between the signals received by the two antennas and defines if the diversity antenna system could properly operate. The ECC can be calculated using either the farfield radiation patterns orS-parameters [33, 34]. However, using S-parameters for the ECC calculation is valid only for the antenna having a very high efficiency [35]. In this antenna design, using S-parameters could cause nonaccurate results particularly around 800 MHz (efficiency around 50%). Therefore, in this study, ECC values are obtained using the farfield radiation patterns. Figure 22 plots the ECC on each LTE frequency bands.Figure 22
ECC value of the MIMO system.The ECC value range is between 0 and 1 with 1 corresponding to similar radiation patterns. The necessary diversity requirement is an ECC value lower than 0.5. For the antenna system, the ECC is lower than 0.25 meaning that the radiation pattern is well reconfigured for the four LTE bands. In [26], authors indicated that the distance between the antenna elements and their respective orientations in the system affect both ∣S21∣ and ECC values. Therefore, the distance of the antenna elements in MIMO system has been optimized, and the antenna orientation is selected to be orthogonal as presented in Figure 12. Finally, a low ECC value is obtained, and the efficiencies of both antenna elements are higher than 50% on all the LTE frequency bands.


## 3.1. Structure of the MIMO Antenna System

The proposed MIMO system integrates two DRAs placed face-to-face in such a way that they can be embedded in the automotive dedicated space. Therefore, the distance between the DRAs is 18 mm along thex-axis and 40 mm for the y-axis. Each DRA is independently fed with a 50 Ω coaxial cable, and the system shape and element locations are symmetric as presented in Figure 17.Figure 17
MIMO antenna system in CST and prototype.


## 3.2. Performance of the MIMO Antenna System

S-parameters are presented in Figures 18(a) and 18(b). Since DRAs are similar and the structure is symmetric, the ∣S11∣ and ∣S22∣ parameters are almost similar, and they are well matched on the four LTE frequency bands. The system reciprocity implies that the ∣S21∣ parameter is equal to the ∣S12∣ parameter. Since the distance between the two radiating elements equals λ0/8.5 at 800 MHz, the coupling is reaching its maximum of −6 dB on the first working band. This coupling will therefore result in a lower realized gain and thus a lower efficiency compared to the antenna studied alone.Figure 18
MIMO system: (a) measured and simulated∣S11∣ and ∣S22∣ parameters and (b) measured and simulated ∣S21∣ parameters.
(a)
(b)Figure19 plots the total efficiency and maximum realized gain according to the frequency. As expected, it demonstrates that the antenna efficiency is lower for frequency bands where the coupling is higher than 10 dB. However, this efficiency remains more than acceptable since it is higher than 50% around 800 MHz, higher than 70% between 1.7 GHz and 2.2 GHz, and better than 60% for the highest LTE frequency band.Figure 19
Measured and simulated maximum realized gains and total efficiencies of the MIMO system.Figure20 shows the 3D measured radiation patterns for the main and the MIMO antennas at 820 MHz, 1575 MHz, 1800 MHz, 2150 MHz, and 2600 MHz. It can be observed that this device offers the possibility to perform a pattern diversity since the radiation pattern is depending on the excited DRA. Figure 21 shows the 2D radiation patterns in the x0y plane (θ=90°). There is a good agreement between the measurement and the simulation.Figure 20
Radiation patterns of the MIMO system in 3D measurement.Figure 21
2D radiation patterns of each DRA elements in the MIMO system.However, these results are not sufficient to conclude that the radiation pattern is reconfigured. An important parameter to evaluate the diversity performances of a system is the envelope correlation coefficient (ECC). Indeed, it evaluates the correlation between the signals received by the two antennas and defines if the diversity antenna system could properly operate. The ECC can be calculated using either the farfield radiation patterns orS-parameters [33, 34]. However, using S-parameters for the ECC calculation is valid only for the antenna having a very high efficiency [35]. In this antenna design, using S-parameters could cause nonaccurate results particularly around 800 MHz (efficiency around 50%). Therefore, in this study, ECC values are obtained using the farfield radiation patterns. Figure 22 plots the ECC on each LTE frequency bands.Figure 22
ECC value of the MIMO system.The ECC value range is between 0 and 1 with 1 corresponding to similar radiation patterns. The necessary diversity requirement is an ECC value lower than 0.5. For the antenna system, the ECC is lower than 0.25 meaning that the radiation pattern is well reconfigured for the four LTE bands. In [26], authors indicated that the distance between the antenna elements and their respective orientations in the system affect both ∣S21∣ and ECC values. Therefore, the distance of the antenna elements in MIMO system has been optimized, and the antenna orientation is selected to be orthogonal as presented in Figure 12. Finally, a low ECC value is obtained, and the efficiencies of both antenna elements are higher than 50% on all the LTE frequency bands.


## 4. Conclusions

A compact and multiband MIMO system has been designed for automotive applications. The proposed antenna is exhibiting all the advantages of using a dielectric resonator antenna by covering LTE 800DD, LTE 1800+, LTE 2100, and LTE 2600. The DRA design satisfies the −6 dB impedance bandwidth for the four distinct frequency bands. A total efficiency better than 50% is achieved in the LTE 800DD band and 75% on the other frequency bands while being miniature. It has global dimensions ofλ0/8×λ0/17×λ0/29 at 790 MHz.Then a MIMO system has been proposed and is composed by two DRAs placed within a small area. It is presenting a low ECC value (under 0.25) while having good radiating performances. All the measurements are well matched with the simulations. Moreover, the proposed antenna has a simple rectangular shape with only one surface printed with copper which can be easily fabricated. With these features, this antenna is very suitable for practical automotive applications.



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*Source: 8231081-2018-12-27.xml*