# A Bipolar Output Active-Switched-Inductor Converter for Bipolar DC Microgrid

Authors: Yingjun Guo; Hexu Sun
Journal: International Journal of Photoenergy
 (2022)

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## Abstract

A bipolar DC microgrid shows good performance since it improves the efficiency and system’s reliability. To transmit power to the load, several DC-DC converters are required by the conventional bipolar DC microgrid, which results into large weight, volume, and high cost. To integrate bipolar DC microgrid with the photovoltaic (PV) module, a novel bipolar output converter is proposed in this paper. This converter possesses very simple circuit configuration and can achieve a high step-up voltage gain by adopting active-switched-inductor (ASL) network; meanwhile, bipolar output voltage is between positive line (P), negative line (N), and common ground (O), which can offer three voltage classes (Vdc, +0.5 Vdc and -0.5 Vdc). To verify the precision of the analysis of the proposed converter, a prototype has been designed and experiments are performed in the lab.


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## Body

## 1. Introduction

As the global energy crisis and environmental pollution become more and more serious, distributed generations such as photovoltaic (PV) and wind power have attracted more and more attention because of its renewable, clean, and other advantages. However, distributed power generation has some disadvantages, such as randomness and intermittent fluctuations. Therefore, connecting new energy to the grid through the microgrid has become a future development trend [1–5]. The microgrid system includes DGs, energy storage units, distributed loads, and interface converters. There are two types of microgrids: DC microgrid and AC microgrid. Since grid exists in the form of communication, most of the current research on microgrid focuses on micro networks. The output of photovoltaic and energy storage units is in the form of DC, and a large number of loads also obtain electrical energy in the form of DC. In this case, the DC microgrid has the following advantages as compared with the AC microgrid [6–8]:
(1)
There are fewer energy conversion stages, which means power losses are reduced and system efficiency is improved(2)
There are fewer power electronic devices; the costs get reduced(3)
The power outputs do not need to be synchronized; the control strategy is simple without considering such factors as frequency, phase angle, and reactive power(4)
The load power supply is not affected by the network voltage drop, sudden rise, phases imbalance, and harmonics wave, so the power quality is betterTherefore, DC microgrids are especially researched all over the world.In the DC microgrid, renewable energy and energy generated by energy storage devices can be transmitted through a bipolar bus or a unipolar bus. DC microgrid can be divided into unipolar microgrid and bipolar microgrid. Compared with the unipolar DC microgrid, the bipolar DC microgrid has the following advantages [9–12]:
(1)
When either bus gets failed, the other buses can work normally(2)
The current through ground is closed to zero, during the normal operation(3)
As compared to the unipolar DC microgrid, the current in the bipolar DC microgrid must be smaller, when the same power is transmitting in the bipolar DC microgrid(4)
Three voltage levels are introduced by the bipolar DC bus, comprising a negative voltage and two symmetrical positive voltageHence, high efficiency, flexibility, and reliability can be achieved by the bipolar DC microgrid system. The structure of the bipolar power grid is shown in Figure1. According to its characteristics, many researches about bipolar output converters have been done [13–15]; Figure 1 illustrates the structure of bipolar DC microgrid, which helps to reduce energy conversion stages and increase microgrid system efficiency. The main problem of bipolar output converter is that different loads will cause bipolar output to be different; in order to solve this problem, some control strategies on voltage balancer have been proposed [16–18].Figure 1
Structure of the bipolar DC microgrid.The voltage stress of the semiconductor devices and the power quality get affected by the unbalance voltage; therefore, it is essential to have symmetrical bipolar output voltage for the application of bipolar DC microgrid. In recent years, the research of voltage balancers is getting further due to the quick development of DC microgrid technologies. The voltage balancer can be used into two applications: isolated and nonisolated. The isolated voltage balancers of DAB are introduced in [19, 20]. However, the isolated voltage balancers require more devices and magnetic components. For some situations where there is no need to provide isolation between the AC and DC grids, nonisolated voltage balancers are needed. Some voltage balancer of buck/boost type applied in bipolar DC bus is proposed in [21–23], and it has been successfully employed to DC microgrid. A voltage balancer employing dual-buck topologies is introduced in [24]. Some three-level (TL) converters with bipolar output are proposed in [25–27], which adopt balance control. In [18], a number of bipolar output converters, based on zeta converter, SEPIC converter, Cuk converter, and buck/boost converter, are introduced that could supply bipolar voltage to the loads. In [28], a bipolar output converter based on a combination of both SEPIC and Cuk converter is proposed, which has no balance control and inherently bipolar output; however, the complexity of converter and a number of components make the efficiency performance not well. Using independent neutral modules is another method to balance the positive and negative DC bus voltages for two-level or three-level converters. In [29], an independently controlled neutral module (ICNM) was used in three-phase four-wire inverter to provide the path for neutral current under unbalanced load condition, which can help balance the positive and negative voltages; however, the resonance will occur between the neutral inductor and the DC-link capacitors, so the neutral inductor current feedback scheme (ICFS) was proposed in [29] to improve the phase margin to ensure system stability. Also, to reduce the neutral point voltage ripple, three improved control schemes, namely, the neutral current feed-forward plus inductor current feedback scheme (NFICFS), proportional resonant cascaded regulator plus inductor current feedback scheme (PR+ICFS), and neutral current feed-forward plus inductor current feedback scheme (NCFF+ICFS), were proposed in [29] to reduce the neutral point voltage ripple and balance the positive and negative DC bus voltages. In [30], power decoupling network (PDN) is embedded for reducing the split DC bus capacitance. To restrain the PDN from producing excessive power losses, partial power processing is proposed, such that the PDN and the split DC bus capacitors share the midpoint current and the neutral current per an optimal current distribution factor.Based on the structure and characteristics of bipolar DC microgrid, this paper suggests a bipolar output DC-DC converter based on ASL; the ASL network has a simple topological structure, the operation mode of the two parallel charging series inductance discharge with high voltage gain, components of current and voltage stress are relatively low; in addition, owning to inherent structure of the suggested converter, balance control is not required. The external characteristics and under various working modes the operation analysis of the converter have been discussed in detail. Finally, to verify the precision of the analysis, a prototype has been designed in the lab and results of experiments and simulations are given.


## 2. Theory of Operation

Figure2 shows the proposed converter with ASL network. The ASL is made up of two inductors and two switches. When the switches are turned on simultaneously, the two inductors are charging in parallel; when the switches are turned off, the inductors are discharging in series, which helps to increase the voltage ratio. The voltage of S2 was clamped by Co1,Ci2, and the voltage of S1 was clamped by Co2,Ci1. And due to the symmetry of the structure, the converter can achieve voltage balance of bipolar output inherently without balance control. In order to analyze the discontinuous conduction mode (DCM) and continuous conduction mode (CCM) of the proposed converter, few assumptions are made.
(1)
Ideal components are used in the circuit, and impact of ESR of capacitance and inductance, forward voltage drops across diodes, and switch’s ON state resistance are ignored(2)
Identical input capacitors, output capacitors, and the inductors are chosen and possess the same values; therefore,(1)L1=L2=L,Ci1=Ci2,Co1=Co2.(3)
The DC loads between PO and NO are symmetrical due to the control of the voltage balancer in DC microgrid, which is the most important for operation of the proposed converter(4)
Assuming the converter and DC microgrid operates in the steady stateFigure 2
The proposed converter.


### 2.1. Analysis of CCM Operation

During the CCM, Figure3 describes the typical characteristics of the proposed converter, with respect to the equivalent circuits and key waveforms illustrated in Figures 3 and 4; the CCM operation mode is analyzed as follows.
(1)
Mode 1t0,t1: the proposed converter equivalent power circuit in mode 1 is illustrated in Figure 4(a) in which switches S1 and S2 are turned ON. In this mode, the inductors are charging in parallel by the input voltage Vi, and the current through inductors increases linearly; the voltage across the inductors can be expressed as(2)L1diL1dt=L2diL2dt=VL1=VL2=Vi.(2)
Mode 2t1,t2: the proposed converter equivalent power circuit in mode 2 is illustrated in Figure 4(b) in which switches S1 and S2 are turned OFF. During this mode, both inductors are discharging in series and the two loads are absorbing power from each of the inductors. The voltage across the inductors is(3)VL1=VCi1−VCo1,VL2=VCi2−VCo2.Figure 3
Key waveform in CCM operation.Figure 4
The equivalent circuit in CCM operation: (a) mode 1 and (b) mode 2.
(a)(b)


### 2.2. Analysis of DCM Operation

The discontinuous conduction mode of the proposed converter is classified into three modes. For discontinuous conduction mode, typical characteristics of the proposed converter are shown in Figure5. Mode 1 and mode 2 are similar as continuous conduction mode; in inductors, no current is flowing after tD is the main difference as shown in Figure 6; in this mode, the loads are only getting charged by VCO1 and VCO2. It can be observed that from time tD to Ts the current of the inductors is zero.Figure 5
Mode 3 in DCM operation.Figure 6
Key waveform in the DCM.


## 2.1. Analysis of CCM Operation

During the CCM, Figure3 describes the typical characteristics of the proposed converter, with respect to the equivalent circuits and key waveforms illustrated in Figures 3 and 4; the CCM operation mode is analyzed as follows.
(1)
Mode 1t0,t1: the proposed converter equivalent power circuit in mode 1 is illustrated in Figure 4(a) in which switches S1 and S2 are turned ON. In this mode, the inductors are charging in parallel by the input voltage Vi, and the current through inductors increases linearly; the voltage across the inductors can be expressed as(2)L1diL1dt=L2diL2dt=VL1=VL2=Vi.(2)
Mode 2t1,t2: the proposed converter equivalent power circuit in mode 2 is illustrated in Figure 4(b) in which switches S1 and S2 are turned OFF. During this mode, both inductors are discharging in series and the two loads are absorbing power from each of the inductors. The voltage across the inductors is(3)VL1=VCi1−VCo1,VL2=VCi2−VCo2.Figure 3
Key waveform in CCM operation.Figure 4
The equivalent circuit in CCM operation: (a) mode 1 and (b) mode 2.
(a)(b)


## 2.2. Analysis of DCM Operation

The discontinuous conduction mode of the proposed converter is classified into three modes. For discontinuous conduction mode, typical characteristics of the proposed converter are shown in Figure5. Mode 1 and mode 2 are similar as continuous conduction mode; in inductors, no current is flowing after tD is the main difference as shown in Figure 6; in this mode, the loads are only getting charged by VCO1 and VCO2. It can be observed that from time tD to Ts the current of the inductors is zero.Figure 5
Mode 3 in DCM operation.Figure 6
Key waveform in the DCM.


## 3. Circuit Performance Analysis

Due to the circuit symmetry of the proposed converter, the performance analysis of the circuit is illustrated as follows.


### 3.1. Voltage Conversion

In order to simplify the analysis, assumptions are as follows: output voltage and input voltage are constant and capacitorsCi1, Ci2, Co1, and Co2 are large enough, so VCi1=Vi/2=VCi2,VCo1=Vo/2=VCo2.By using voltage-second balance on inductorsL1,L2,
(4)D⋅Vi+1−D⋅VCi2−VCo2=0,(5)D⋅Vi+1−D⋅VCi2−VCo2=0.Simplifying (4) and (5), the voltage gain of the CCM operation is
(6)GCCM=VoVi=D+11−D.


### 3.2. DCM Voltage Conversion Ratio

Under DCM operation, the inductor’s current increases, when the switches are conducting and its peak current is(7)iL1p=iL2p=ViDTsL.The inductor’s current decreases, when the switches are off, and its peak current is(8)iL1p=iL2p=Vo−ViD2Ts2L,whereD2Ts is the duration time of mode 2.D2 can be derived from (8) and (7) as follows:
(9)D2=2ViVo−ViD.The average current of output capacitor in a period is zero, so(10)Io=iL1pD22=VoRL.Substituting (9) into (10), GDCM is derived as
(11)GDCM=12+14+D2τ,whereτ is a time constant which can be defined as τ=Lfs/RL.The voltage gain of the CCM is equal to that of the DCM when the converter is operating in critical continuous state; according to (6) and (11), intrinsic time τ can be derived as
(12)τ′=D1−D221+D.According to (8) and (9), in the condition of critical continuous state, the current of loads is
(13)Io′=ΔIL21−D=VoTsD1−D22L1+D.WhenD=0.281, the current of loads reaches the maximum.
(14)Iomax′=0.0567VoLfs.So, the current of loads can be represented as(15)Io′=Iomax′D1−D20.05671+D.According to (6), (10), and (15), Figure 7 shows the relationship between duty cycle, current of loads, and gain.Figure 7
The relationship between duty cycle, current of loads, and gain.


### 3.3. Efficiency Analysis considering Nonideality under CCM

Figure8 illustrates the equivalent circuit of the proposed converter with nonidealities. It is considered that the inductors are alike and rL is their effective series resistance (L1=L2=L and rL1=rL2=rL). Also, it is considered that all the diodes D are alike and possess forward voltage drop VFD (VFD1=VFD2) and forward resistance rD (rD1=rD2). The two switches are also alike and possess ON state resistance rS (rS1=rS2).Figure 8
The equivalent circuit of the proposed converter with nonidealities.In mode I, the average voltage across the inductors and the average current through the output capacitor are derived as(16)IC1=−Vo1Ro1=−Vo2Ro1,IC2=−Vo2Ro2=−Vo2Ro1,VL1=VL2=Vin−iLrL+rS.In mode II, average voltage across the inductors and the average current through the output capacitors are derived as(17)IC1=−Vo1Ro1=−Vo2Ro1,IC2=−Vo2Ro2=−Vo2Ro2,VL1=VL2=12Vin−iL2rL+2rD−Vo−2VFD.The voltage gain of the converter(18)Greal=1+D−A1−D+B=Gideal−A/1−D1+B/1−D,WhereA=21−DVFD and B=2rL+DrS+1−DrD/Ro1.The proposed converter’s power loss and efficiency can be derived as(19)η′=1−D2−21−D2VFD1−D2+rL+DrS+1−DrD1+D/Ro1.


### 3.4. Voltage and Current Stress of the Power Devices

The diode and power switch current and voltage stresses with respect to the analysis of CCM operation modes are(20)VD1=VD2=VS1=VS2=11+DVo,IS1=IS2=IL+ΔIL=Io+D1−DVoD+12fsL,ID1=ID2=ΔIL+IL=Io+D1−DVoD+12fsL.As shown in Figure9, the voltage stress of all power devices is lower than output voltage, which greatly reduces the power loss of switches and solves the reverse recovery problem of diodes; the current stress of power devices is also reduced due to the ASL unit. In addition, thanks to the clamping structure of four capacitors, not only the circuit resonance problem is well solved when the switch is turned off but also the voltage on two switches and diodes can keep balance.Figure 9
The relationships between voltage stress on switches and diodes with duty cycle.


### 3.5. The Analysis of Input Current and Output Voltage Ripple

According to the current waveform in Figure3, the current through inductors and neutral line can be derived as
(21)iL1=iL2=ILt0+ViLt−t0,t0≤t<t1,ILt1+Vi−Vo2Lt−t2,t1≤t<t2,iM=0.So, the input current can de expressed as(22)iin=2VoR1−D−D1−DVofsL1+D+1−D2VoL1+Dt−t0,t0≤t<t1,VoR1−D+Vo1−DD2fsL1+D+DVoL1+Dt−t2,t1≤t<t2,Δiin=Vo1−DRo+3D1−DVo2fsLD+1.The current effective value of the switches is(23)Irms‐s=1T∫DT0IL−12ΔIL+ΔILDTt2=IoG+1G−12112Ki2+1,whereKi=△IL/iL.The ripple of output capacitor voltage with respect to Figure10 can be derived as
(24)ΔVo1t=ΔVo2t=1C1∫tt0Iodt,t0<t<t1,1C1∫tt1iL1t−Iodt,t1<t<t2,ΔVo=ΔVo1+ΔVo2=VoDRofsCo.Figure 10
The waveforms of output.


## 3.1. Voltage Conversion

In order to simplify the analysis, assumptions are as follows: output voltage and input voltage are constant and capacitorsCi1, Ci2, Co1, and Co2 are large enough, so VCi1=Vi/2=VCi2,VCo1=Vo/2=VCo2.By using voltage-second balance on inductorsL1,L2,
(4)D⋅Vi+1−D⋅VCi2−VCo2=0,(5)D⋅Vi+1−D⋅VCi2−VCo2=0.Simplifying (4) and (5), the voltage gain of the CCM operation is
(6)GCCM=VoVi=D+11−D.


## 3.2. DCM Voltage Conversion Ratio

Under DCM operation, the inductor’s current increases, when the switches are conducting and its peak current is(7)iL1p=iL2p=ViDTsL.The inductor’s current decreases, when the switches are off, and its peak current is(8)iL1p=iL2p=Vo−ViD2Ts2L,whereD2Ts is the duration time of mode 2.D2 can be derived from (8) and (7) as follows:
(9)D2=2ViVo−ViD.The average current of output capacitor in a period is zero, so(10)Io=iL1pD22=VoRL.Substituting (9) into (10), GDCM is derived as
(11)GDCM=12+14+D2τ,whereτ is a time constant which can be defined as τ=Lfs/RL.The voltage gain of the CCM is equal to that of the DCM when the converter is operating in critical continuous state; according to (6) and (11), intrinsic time τ can be derived as
(12)τ′=D1−D221+D.According to (8) and (9), in the condition of critical continuous state, the current of loads is
(13)Io′=ΔIL21−D=VoTsD1−D22L1+D.WhenD=0.281, the current of loads reaches the maximum.
(14)Iomax′=0.0567VoLfs.So, the current of loads can be represented as(15)Io′=Iomax′D1−D20.05671+D.According to (6), (10), and (15), Figure 7 shows the relationship between duty cycle, current of loads, and gain.Figure 7
The relationship between duty cycle, current of loads, and gain.


## 3.3. Efficiency Analysis considering Nonideality under CCM

Figure8 illustrates the equivalent circuit of the proposed converter with nonidealities. It is considered that the inductors are alike and rL is their effective series resistance (L1=L2=L and rL1=rL2=rL). Also, it is considered that all the diodes D are alike and possess forward voltage drop VFD (VFD1=VFD2) and forward resistance rD (rD1=rD2). The two switches are also alike and possess ON state resistance rS (rS1=rS2).Figure 8
The equivalent circuit of the proposed converter with nonidealities.In mode I, the average voltage across the inductors and the average current through the output capacitor are derived as(16)IC1=−Vo1Ro1=−Vo2Ro1,IC2=−Vo2Ro2=−Vo2Ro1,VL1=VL2=Vin−iLrL+rS.In mode II, average voltage across the inductors and the average current through the output capacitors are derived as(17)IC1=−Vo1Ro1=−Vo2Ro1,IC2=−Vo2Ro2=−Vo2Ro2,VL1=VL2=12Vin−iL2rL+2rD−Vo−2VFD.The voltage gain of the converter(18)Greal=1+D−A1−D+B=Gideal−A/1−D1+B/1−D,WhereA=21−DVFD and B=2rL+DrS+1−DrD/Ro1.The proposed converter’s power loss and efficiency can be derived as(19)η′=1−D2−21−D2VFD1−D2+rL+DrS+1−DrD1+D/Ro1.


## 3.4. Voltage and Current Stress of the Power Devices

The diode and power switch current and voltage stresses with respect to the analysis of CCM operation modes are(20)VD1=VD2=VS1=VS2=11+DVo,IS1=IS2=IL+ΔIL=Io+D1−DVoD+12fsL,ID1=ID2=ΔIL+IL=Io+D1−DVoD+12fsL.As shown in Figure9, the voltage stress of all power devices is lower than output voltage, which greatly reduces the power loss of switches and solves the reverse recovery problem of diodes; the current stress of power devices is also reduced due to the ASL unit. In addition, thanks to the clamping structure of four capacitors, not only the circuit resonance problem is well solved when the switch is turned off but also the voltage on two switches and diodes can keep balance.Figure 9
The relationships between voltage stress on switches and diodes with duty cycle.


## 3.5. The Analysis of Input Current and Output Voltage Ripple

According to the current waveform in Figure3, the current through inductors and neutral line can be derived as
(21)iL1=iL2=ILt0+ViLt−t0,t0≤t<t1,ILt1+Vi−Vo2Lt−t2,t1≤t<t2,iM=0.So, the input current can de expressed as(22)iin=2VoR1−D−D1−DVofsL1+D+1−D2VoL1+Dt−t0,t0≤t<t1,VoR1−D+Vo1−DD2fsL1+D+DVoL1+Dt−t2,t1≤t<t2,Δiin=Vo1−DRo+3D1−DVo2fsLD+1.The current effective value of the switches is(23)Irms‐s=1T∫DT0IL−12ΔIL+ΔILDTt2=IoG+1G−12112Ki2+1,whereKi=△IL/iL.The ripple of output capacitor voltage with respect to Figure10 can be derived as
(24)ΔVo1t=ΔVo2t=1C1∫tt0Iodt,t0<t<t1,1C1∫tt1iL1t−Iodt,t1<t<t2,ΔVo=ΔVo1+ΔVo2=VoDRofsCo.Figure 10
The waveforms of output.


## 4. Design of Parameters

In order to verify the analysis, we have established a prototype in laboratory; the following is the main design criterion.


### 4.1. Inductor Design

The same equation can be used to calculate the current ripple of the inductorsL1 and L2. The inductor’s section depends on duty cycle (D), inductor ripple current (△iL), inductor voltage (VL), and switching frequency (fs). Vi is equal to the voltage of the input inductance L1 and L2. Hence, the inductance value of L1 and L2 can be expressed as
(25)L1=L2≥ViDTsKiiL.


### 4.2. Capacitor Design

The value of the output capacitorsCo1 and Co2 depends on the switching frequency fs, output voltage Vo, converter’s output power Po, and voltage ripple △Vo. The rating of the capacitor C can be obtained as follows:
(26)Co1=Co2≥21−DTsIoKVVo,whereKV=△Vo/Vo.


### 4.3. Power Device Design

According to the working condition and formulas abovementioned, the maximum of voltage stress of all power devices is 160 V, the theory current stress of switches is 2 A, and the theory current stress of diodes is 0.8 A. So, following these data and considering margin setting, the specific device selection can be easily found.


### 4.4. Controller Design Consideration

In DC microgrid, the second stage of the step-up DC-DC converter is usually a single-phase inverter load; the second harmonic current (SHC) will occur in the DC-DC converter when the instantaneous input and output power of two-stage single-phase converter is imbalanced. The SHC will propagate back into the front-end DC-DC converter and the input DC voltage source. For the front-end DC-DC converter, the SHC will lead to higher current stress for power switches and produce extra power losses in power switches and magnetic components. For the input DC voltage sources, such as batteries or fuel cells, the SHC will shorten the lifetime. To reduce the SHC, in [31, 32], a virtual-impedance approach is proposed; the virtual series impedance is realized by the feedback of the boost-diode current or the boost-inductor current, while the virtual parallel impedance is implemented by the feedback of the DC bus voltage. Based on the approach, the SHC reduction control schemes are proposed. The proposed converter is similar to the prestage boost converter in [31, 32], so the controller design of the proposed converter can refer to the closed-loop control scheme in these two papers.


## 4.1. Inductor Design

The same equation can be used to calculate the current ripple of the inductorsL1 and L2. The inductor’s section depends on duty cycle (D), inductor ripple current (△iL), inductor voltage (VL), and switching frequency (fs). Vi is equal to the voltage of the input inductance L1 and L2. Hence, the inductance value of L1 and L2 can be expressed as
(25)L1=L2≥ViDTsKiiL.


## 4.2. Capacitor Design

The value of the output capacitorsCo1 and Co2 depends on the switching frequency fs, output voltage Vo, converter’s output power Po, and voltage ripple △Vo. The rating of the capacitor C can be obtained as follows:
(26)Co1=Co2≥21−DTsIoKVVo,whereKV=△Vo/Vo.


## 4.3. Power Device Design

According to the working condition and formulas abovementioned, the maximum of voltage stress of all power devices is 160 V, the theory current stress of switches is 2 A, and the theory current stress of diodes is 0.8 A. So, following these data and considering margin setting, the specific device selection can be easily found.


## 4.4. Controller Design Consideration

In DC microgrid, the second stage of the step-up DC-DC converter is usually a single-phase inverter load; the second harmonic current (SHC) will occur in the DC-DC converter when the instantaneous input and output power of two-stage single-phase converter is imbalanced. The SHC will propagate back into the front-end DC-DC converter and the input DC voltage source. For the front-end DC-DC converter, the SHC will lead to higher current stress for power switches and produce extra power losses in power switches and magnetic components. For the input DC voltage sources, such as batteries or fuel cells, the SHC will shorten the lifetime. To reduce the SHC, in [31, 32], a virtual-impedance approach is proposed; the virtual series impedance is realized by the feedback of the boost-diode current or the boost-inductor current, while the virtual parallel impedance is implemented by the feedback of the DC bus voltage. Based on the approach, the SHC reduction control schemes are proposed. The proposed converter is similar to the prestage boost converter in [31, 32], so the controller design of the proposed converter can refer to the closed-loop control scheme in these two papers.


## 5. Simulation and Experiment Result

To validate the accuracy of the theoretical analysis, the proposed converter prototype is designed and experiments are performed in the lab, and experimental conditions are given in Table1. According to the last section, the components and parameters are summarized in Table 2.Table 1
Experimental conditions.
ComponentsParametersVi (input voltage)60-80 VVo (output voltage)±120 VPo (rated power)200 Wfs (switching frequency)50 kHzTable 2
Experimental conditions.
ComponentsParametersS1,S2 (switches)IRFP250D1,D2 (diodes)SF24L1,L2 (inductors)240μHCi1,Ci2 (input capacitors)1000μF/100 VCo1,Co2 (output capacitors)470μF/250 V


### 5.1. Simulation and Experimental Results

In the ideal condition, the simulation of the proposed converter is designed on Saber. As shown in Figures11 and 12, the simulation and experimental results of the proposed converter are given to check analysis; it is obvious that experimental results are consistent with simulation results.Figure 11
Simulation waveforms. (a)Vgs1,Vgs2,Vds1,Vds2; (b) Vgs1,Vgs2,VD1,VD2; (c) Vgs1,Vgs2,iL1,iL2; (d) Vin,Vo1,Vo2.
(a)(b)(c)(d)Figure 12
Experiment waveforms. (a)Vgs1,Vgs2,Vds1,Vds2; (b) Vgs1,Vgs2,VD1,VD2; (c) Vgs1,Vgs2,iL1,iL2; (d) Vin,Vo1,Vo2.
(a)(b)(c)(d)The converter is tested for output power 200 W (loadR=290Ω), input voltage 60 V, and duty cycles D=60%. The voltage waveforms of switches S1,S2 are shown in Figures 11(a) and 12(a), the voltage stress VS1,VS2 is equal to theoretical half of the sum of input voltage and output voltage when the inductors are discharging, which benefit from the voltage-clamping topology of all capacitors. As shown in Figures 11(b) and 12(b), similar to switches, the voltage stresses of output diodes are also clamped by capacitors when switches are on and the voltage stress is the same as that of switches. Figures 11(c) and 12(c) show the inductor’s current waveforms; the current is increasing linearly when switches are on and decreasing linearly when switches are off, which are consistent with the theoretical analysis. As for bipolar output converter designed for bipolar DC microgrid, the typical character is shown in Figures 11(d) and 12(d); thanks to the inherent symmetric structure, the voltage on two loads is the same and bipolar, which means Vo1=+120V and Vo2=−120V over the ground. Figure 13 shows sudden load change experiment of the proposed converter. According to Table 2, the rated power of the proposed converter is 200 W. The sudden load increase experiment is shown in Figure 13(a), which is from 200 W to 100 W. Sudden load decrease experiment is shown in Figure 13(b), which is from 100 W to 200 W. As shown in Figure 13(a), the output current decreases from 0.83 A to 0.42 A in a stepwise manner, and the proposed converter can balance the bipolar output voltages under a sudden load increase. As shown in Figure 13(b), the output current increases from 0.42 A to 0.83 A in a stepwise manner, and the proposed converter also can balance the bipolar output voltages under a load sudden decrease.Figure 13
Sudden load change experiments: (a) sudden load increase and (b) sudden load decrease.
(a)(b)


### 5.2. Efficiency and Power Loss

The designed prototype’s efficiency is probed at various input voltage and power levels by experimental measurement. The efficiency plot for various input voltage and power levels is shown in Figure14. It can be seen that the proposed converter shows a better efficiency performance when Po=100W and the efficiency exceeds 97.4% when Vi=80V.Figure 14
Efficiency curves.The experimental conditions are shown in Table2. According to Table 2, the detailed loss analysis is as follows:
(1)
The losses of the switches (S1 and S2) result from switching and conduction, and they can be estimated as follows:(27)PS=2Isrms2RDSon+12fsVsIston+toff=3.325W,where Isrms is the RMS current of the switches and RDSon is the on-state resistance of the switches; fs is switching frequency; ton and toff are turn-on time and turn-off time, respectively; Vs is the voltage stress of the switches, and Is is the current of the switches during switching time
(2)
The diode losses are given by(28)PD=2IDavgvF=1.533W,where IDrms is the average current of the diodes and vF is the diode forward voltage
(3)
The inductor losses are given by(29)PL=2rLILrms2+aBpkbfscAele=1.112W,where ILrms is the RMS current of the inductor; rL is copper resistance; Bpk is the AC magnetic flux density for the magnetic device core; a, b, and c are constants determined from the curve fitting of the core; le is the core medium path length; and Ae is the transversal core area
(4)
The capacitor losses can be given by(30)PC=2ICrms2ESR=0.22W,where ICrms is the RMS current of the capacitor and ESR is the resistor of capacitors.So, the loss distribution of the proposed converter for working condition ofPo=200W, fs=50kHz, Vi=60V, and Vo=240V is shown in Table 3 and Figure 15.Table 3
Loss distribution.
ComponentsLosses (W)PercentageSwitches3.325 W53.8%Diodes1.533 W24.8%Inductors1.112 W18%Capacitors0.22 W3.4%Figure 15
Power loss distribution.


## 5.1. Simulation and Experimental Results

In the ideal condition, the simulation of the proposed converter is designed on Saber. As shown in Figures11 and 12, the simulation and experimental results of the proposed converter are given to check analysis; it is obvious that experimental results are consistent with simulation results.Figure 11
Simulation waveforms. (a)Vgs1,Vgs2,Vds1,Vds2; (b) Vgs1,Vgs2,VD1,VD2; (c) Vgs1,Vgs2,iL1,iL2; (d) Vin,Vo1,Vo2.
(a)(b)(c)(d)Figure 12
Experiment waveforms. (a)Vgs1,Vgs2,Vds1,Vds2; (b) Vgs1,Vgs2,VD1,VD2; (c) Vgs1,Vgs2,iL1,iL2; (d) Vin,Vo1,Vo2.
(a)(b)(c)(d)The converter is tested for output power 200 W (loadR=290Ω), input voltage 60 V, and duty cycles D=60%. The voltage waveforms of switches S1,S2 are shown in Figures 11(a) and 12(a), the voltage stress VS1,VS2 is equal to theoretical half of the sum of input voltage and output voltage when the inductors are discharging, which benefit from the voltage-clamping topology of all capacitors. As shown in Figures 11(b) and 12(b), similar to switches, the voltage stresses of output diodes are also clamped by capacitors when switches are on and the voltage stress is the same as that of switches. Figures 11(c) and 12(c) show the inductor’s current waveforms; the current is increasing linearly when switches are on and decreasing linearly when switches are off, which are consistent with the theoretical analysis. As for bipolar output converter designed for bipolar DC microgrid, the typical character is shown in Figures 11(d) and 12(d); thanks to the inherent symmetric structure, the voltage on two loads is the same and bipolar, which means Vo1=+120V and Vo2=−120V over the ground. Figure 13 shows sudden load change experiment of the proposed converter. According to Table 2, the rated power of the proposed converter is 200 W. The sudden load increase experiment is shown in Figure 13(a), which is from 200 W to 100 W. Sudden load decrease experiment is shown in Figure 13(b), which is from 100 W to 200 W. As shown in Figure 13(a), the output current decreases from 0.83 A to 0.42 A in a stepwise manner, and the proposed converter can balance the bipolar output voltages under a sudden load increase. As shown in Figure 13(b), the output current increases from 0.42 A to 0.83 A in a stepwise manner, and the proposed converter also can balance the bipolar output voltages under a load sudden decrease.Figure 13
Sudden load change experiments: (a) sudden load increase and (b) sudden load decrease.
(a)(b)


## 5.2. Efficiency and Power Loss

The designed prototype’s efficiency is probed at various input voltage and power levels by experimental measurement. The efficiency plot for various input voltage and power levels is shown in Figure14. It can be seen that the proposed converter shows a better efficiency performance when Po=100W and the efficiency exceeds 97.4% when Vi=80V.Figure 14
Efficiency curves.The experimental conditions are shown in Table2. According to Table 2, the detailed loss analysis is as follows:
(1)
The losses of the switches (S1 and S2) result from switching and conduction, and they can be estimated as follows:(27)PS=2Isrms2RDSon+12fsVsIston+toff=3.325W,where Isrms is the RMS current of the switches and RDSon is the on-state resistance of the switches; fs is switching frequency; ton and toff are turn-on time and turn-off time, respectively; Vs is the voltage stress of the switches, and Is is the current of the switches during switching time
(2)
The diode losses are given by(28)PD=2IDavgvF=1.533W,where IDrms is the average current of the diodes and vF is the diode forward voltage
(3)
The inductor losses are given by(29)PL=2rLILrms2+aBpkbfscAele=1.112W,where ILrms is the RMS current of the inductor; rL is copper resistance; Bpk is the AC magnetic flux density for the magnetic device core; a, b, and c are constants determined from the curve fitting of the core; le is the core medium path length; and Ae is the transversal core area
(4)
The capacitor losses can be given by(30)PC=2ICrms2ESR=0.22W,where ICrms is the RMS current of the capacitor and ESR is the resistor of capacitors.So, the loss distribution of the proposed converter for working condition ofPo=200W, fs=50kHz, Vi=60V, and Vo=240V is shown in Table 3 and Figure 15.Table 3
Loss distribution.
ComponentsLosses (W)PercentageSwitches3.325 W53.8%Diodes1.533 W24.8%Inductors1.112 W18%Capacitors0.22 W3.4%Figure 15
Power loss distribution.


## 6. Conclusions

In this paper, a novel bipolar output active-switched-inductor converter was proposed for bipolar DC microgrid. The proposed converter was analyzed from CCM operation analysis, DCM operation analysis, voltage conversion, efficiency analysis considering nonideality, voltage and current stress of power devices, input and output ripple analysis, and design consideration. Finally, the correctness of the theoretical analysis was verified by the results of the simulation and experiment.The proposed converter has the advantages of high voltage gain and wide duty range, and the operation principle is also simple. The voltage and current stress of power device is relatively low due to the active-switched-inductor network. The proposed converter can output bipolar voltage; the bipolar output voltage is between positive line (P), negative line (N), and common ground (O), which can offer three voltage classes (Vdc, +0.5 Vdc and -0.5 Vdc). Also, due to the symmetry of the structure, the proposed converter can achieve voltage balance of bipolar output with easy control. The highest efficiency of the proposed converter is about 97.4%.



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*Source: 7252163-2022-01-19.xml*