# A 25 kW, 25 kHz Induction Heating Power Supply for MOVPE System Using L-LC Resonant Inverter

Authors: Mangesh Borage; Sunil Tiwari
Journal: Advances in Power Electronics
 (2013)

---

## Abstract

A topology named L-LC resonant inverter (RI) for induction heating (IH) applications takes most of the merits of the conventional series and parallel resonant schemes, while eliminating their limitations. In this paper, fundamental frequency AC analysis of L-LC RI is revisited, and a new operating point is suggested featuring enhanced current gain and near in-phase operation as compared to the conventional operating point. An approximate analysis of the circuit with square-wave voltage source is also described highlighting the effect of auxiliary inductor on the source current waveform. The analysis also leads to an optimum choice of the auxiliary inductance. The requirements of the metal organic vapour phase epitaxy (MOVPE) system in which a graphite susceptor is required to be heated to 1200°C demanding a 25 kW, 25 kHz IH power supply, the configuration of developed IH system, and experimental results are presented.


---

## Body

## 1. Introduction

The induction heating (IH) [1] is commonly used for heat treatment of metals (hardening, tempering, and annealing), heating prior to deformation (forging, swaging, upsetting, bending, and piercing), brazing and soldering, shrink fitting, coating, melting, crystal growing, cap sealing, sintering, carbon vapor deposition, epitaxial deposition, and plasma generation. IH is a noncontact method. The heat is generated only in the part, not in the surrounding area except by radiation. The location of the heating can be defined to a specified area on the metal component, thereby achieving accurate and consistent results. As heating occurs in the object itself, IH is considered more efficient than alternative methods.An IH system comprises a basic induction power source which provides the required power output at the required power frequency, complete with matching components, an induction coil assembly, a method of material handling, and some method of cooling. Generally, full-bridge or half-bridge resonant inverters (RIs) are most commonly used as the power supplies for IH. The equivalent model of an IH coil with work piece can be represented in simplified form by an equivalent inductance (Leq) and resistance (Req) as shown in Figure 1. If the IH coil is directly fed from a supply, the ratio of apparent to real power will be large. Therefore, the IH coil is properly compensated by capacitors and additional inductors in suitable configuration so that minimum reactive power is drawn from the source. Additionally to match the load voltage-current requirements to the available source, a matching network is required. The matching is normally achieved with the help of an isolation transformer of suitable turns ratio.Figure 1
Equivalent circuit representation of an IH coil.Based on the connection of compensating capacitor with the IH coil, the two most commonly used RI topologies are as follows.(1)
Series resonant inverter (SRI): the compensating capacitor is placed in series with the IH coil, and it is fed by a voltage source [2–6].(2)
Parallel resonant inverter (PRI): the compensating capacitor is placed in parallel with the IH coil, and it is fed by the current source [7–11].The analysis of these circuits has been performed in great details, and the comparative assessment is also reported in the literature [12, 13].A topology named L-LC RI has been proposed for IH applications [14–21], which takes most of the merits of SRI and PRI while eliminating their limitations. It operates with input dc voltage source, thereby eliminating bulky input current smoothening inductor. It offers high current gain, which in turn reduces the current rating of the secondary winding of matching transformer and the feeder to coil.Metal organic vapour phase epitaxy (MOVPE) [22] is a highly controlled method for the deposition of semiconductor epitaxial layers and heterostructures, which are required for the development of several optoelectronic and electronic devices. The process of MOVPE involves a vapour phase reaction between metalorganic compound and a hydride gas, which are transported to a heated (around 1200°C) graphite susceptor, resulting in a growth of the desired material. IH is one of the preferred methods for noncontact heating of the susceptor.The paper aims to investigate the characteristics of L-LC RI for an application demanding a 25 kW, 25 kHz IH power supply to heat graphite susceptor to 1200°C in a MOVPE system for the growth of nitride semiconductors being developed at our Institute. Section2 describes AC analysis of L-LC CN and investigates various characteristics when the converter is operated at the resonant frequency. The suggested operating point is different than the previously proposed operating point which provides enhanced current gain with smaller auxiliary inductance and results in near in-phase operation. Section 3 describes the operation of high-Q L-LC compensating network (CN) with square-wave voltage source leading to an optimum choice of the auxiliary inductor. The requirements, description, and design of the practical system are discussed in Section 4. Experimental results are presented in Section 5.


## 2. Analysis of L-LC CN

Figure2 shows the L-LC RI. Input dc source Vd could be an unregulated source (obtained with single-phase or three-phase diode rectifier and filter) or could be a regulated voltage source (obtained with single-phase or three-phase diode/SCR rectifier and filter or another front-end switch-mode regulator). In the former case, regulation of the power to the work piece should be done in the RI stage using frequency variation [23, 24], fixed-frequency pulse width modulation (PWM) [25–27], or pulse density modulation (PDM) [28, 29]. In the latter case, the output control can be done by varying Vd, providing easy control of the output power over a wide range. However, two cascaded converters tend to reduce the overall efficiency.Figure 2
Circuit diagram of L-LC RI for IH application.The following analysis based on the fundamental frequency approximation examines the important characteristics of L-LC topology. In an equivalent circuit of L-LC CN, shown in Figure3, input voltage source is assumed to be a sinusoidal voltage source whose rms value is equal to the rms value of the fundamental component of the square-wave excitation. Following definitions are made for the analysis.Figure 3
Equivalent circuit diagram of L-LC RI for analysis.Angular resonant frequency:(1)ωo=2πfo=1LeqC.Normalized operating frequency:(2)ωn=ωωo,
where ω=2πf is the angular operating frequency and f is  the operating or switching frequency.Characteristic Impedance:(3)Zn=LeqC.CircuitQ:
(4)Q=ωoLeqReq=ZnReq.Inductance ratio:(5)γ=LaLeq.The expression for normalized current in coil and normalized source current (or the current in inductorLa) can be, respectively, derived as
(6)ILeqN=ILeqVd/Zn=22π1(1/Q)(1-ωn2γ)+j[ωn(1+γ)-ωn3γ],ILaN=ILaVd/Zn=22π(1-ωn2)+jωn(1/Q)(1/Q)(1-ωn2γ)+j[ωn(1+γ)-ωn3γ].Next, the salient characteristics of the converter operating at the proposed operating point,ωn=1, are investigated, leading to the following observations:(
1
)
(7)
|
I
L
eq
N
|
ω
n
=
1
=
2
2
π
Q
Q
2
+
(
1
-
γ
)
2
.
For Q≫|(1-γ)|,
(8)|ILeqN|ωn=1≈22π,(
2
)
(9)
|
I
L
a
N
|
ω
n
=
1
=
2
2
π
1
Q
2
+
(
1
-
γ
)
2
,
(10)
ϕ
|
ω
n
=
1
=
∠
I
L
a
N
|
ω
n
=
1
=
-
ta
n
-
1
(
γ
-
1
Q
)
.
The source current at ωn=1 is also seen to be inductive for all values of γ>1, and the phase angle is function of Q and γ. If γ=1, the source current is always in-phase with voltage.Figure4(a) shows the plot of ILeqN as a function of γ for different values of Q. When the converter is operated at ωn=1, ILeqN is seen to be relatively insensitive to γ, particularly at high values of Q—a typical operating condition in an IH application.The plots ofILeqN (a) and ϕ (b) as a function of γ for different values of Q at two operating points.
(a)
(b)Figure4(b) shows the plot of ϕ as a function of γ. The source current lags the applied square-wave voltage ωn=1. While this lagging current is beneficial for zero-voltage-switching (ZVS) of the semiconductor switches, it can be anticipated that a higher value of ϕ results in higher source current causing more conduction loss in the switches. It is evident from the plots of Figure 4(b) that ϕ is small for operation at ωn=1, particularly for operation with high Q. Therefore, the source current and conduction loss in the switches will be smaller at this operating point.The current gain of the CN,H, is given by
(11)H=ILeqNILaN=1(1-ωn2)+jωn(1/Q),
which can further be simplified at ωn=1 as
(12)|H|ωn=1=Q.
Figure 5 shows the plot of H as a function of ωn for γ=5 and different values of Q. Maximum current gain (equal to Q) is observed for operation at the proposed operating point ωn=1.Figure 5
The plots ofH as a function of ωn for γ=5.The characteristics and design of L-LC RI for IH applications operating atωn=ωo_c=(1+γ)/γ have been reported extensively in the literature [15–21]. At this operating point,
(13)|H|ωn=ωo_c=QγQ2+γ(1+γ).
For Q≫γ, |H|ωn=ωo_c≈γ, which is less than theoretical maximum value given by (12) at the proposed operating point. Thus, as also shown in Figure 5, the proposed operating point of L-LC RI results in the enhanced current gain. Further, the expression for ϕ at the conventional operating point can be written as
(14)ϕ|ωn=ωo_c=tan-1(-γQ1+γγ).
The plots of (14) as a function of γ are shown in Figure 4(b) for Q=5 and Q=20. It can be seen that phase angle is smaller at the proposed operating point than the conventional operating point, thereby resulting in less conduction loss in the switches.


## 3. Behaviour of L-LC CN with Square-Wave Voltage Source

Analysis based on the fundamental frequency approximation presented in Section2 suggests that the converter, when operated at ωn=1 offers highest current gain and small phase angle between source voltage and current resulting in low reactive power loading and low conduction loss in the semiconductor devices of the square-wave inverter.  The analysis presented in Section 2 assumes sine wave voltage source at the input. In practice, the input voltage is a square- wave generated by operating switches S1–S4 in Figure 2 at 50 per cent duty cycle. In this section, the behaviour of L-LC resonant inverter for operation at ωn=1 with square-wave voltage source examined. Equivalent circuit for L-LC RI for analysis with square-wave voltage source is shown in Figure 6(a). The source voltage vs can be defined as
(15)vs(t)=Vd·sign(sin(ωot)),vs can be decomposed into its fundamental component vs1 and harmonics vh as
(16)vs(t)=vs1(t)+vh(t)=4Vdπsin(ωot)+vh(t).
From (15) and (16)
(17)vh(t)=Vd·sign(sin(ωot))-4Vdπsin(ωot)=Vd-4Vdπsin(ωot)for0≤t≤Ts2=-Vd-4Vdπsin(ωot)forTs2≤t≤Ts,
wherein Ts is the switching period. Thus, equivalent circuit of Figure 6(a) can be redrawn as Figure 6(b), wherein vs decomposed into vs1 and vh.(a) Equivalent circuit for L-LC RI for analysis withvs. (b) Equivalent circuit redrawn with vs decomposed into vs1 and vh. (c) Approximate equivalent circuit for fundamental frequency. (d) Approximate equivalent circuit for harmonics frequencies.
(a)
(b)
(c)
(d)Equivalent impedance (Zeq) of the Leq, Req, and C resonant circuit at ωn=1 is given by
(18)|Zeq|ωn=1=ReqQ1+Q2≈ReqQ2forQ≫1.
Further, for high-Q resonant circuit in IH applications, Zeq can be approximated to be zero for all operating points other than ωn=1. Under this assumption, equivalent circuit for Figure 6(b) can be decoupled for operation at ωn=1 and at harmonic frequencies, as shown in Figures 6(c) and 6(d), respectively. Assuming ReqQ2≫ωoLa, equivalent circuit at ωn=1 can further be simplified as shown in Figure 6(c). With this simplification, the circuit can be analyzed for fundamental and harmonic frequencies separately to find out individual source current and then be added to find out resultant source current. An expression for is can be derived as
(19)is(t)=is1(t)+ih(t),is1(t)=4VdπQ2Reqsin(ωot),ih(t)=VdLa(t-Ts4)-4VdπωoLasin(ωot-π2)2222222222222222222for0≤t≤Ts2=-VdLa(t-3Ts4)-4VdπωoLasin(ωot-π2)222222222222222222222forTs2≤t≤Ts.
Figure 7(a) shows the calculated waveforms for the operation of the circuit under low-Q condition. It can be easily noticed that is1(t), being inversely proportional to Q, is significantly larger than ih(t) for low-Q values. Therefore, resulting is(t) is also nearly sinusoidal. Under this condition, the predictions of the ac analysis (Section 2) are reasonably accurate. However, under high-Q condition, as shown in Figure 7(b), amplitude of is1(t) is much smaller than ih(t). Therefore, is(t) is nearly the same as ih(t) and is highly nonsinusoidal. Under this condition, the predictions of Section 2 tend to be erroneous. Since the analysis of Section 2 does not account for harmonics, the actual peak and rms values of is calculated by (19) are significantly more than those predicted by (9). This will result in degradation of the current gain from its value predicted by (12). However, the amplitude and rms value of ih(t) can be controlled by choosing a proper value of La. Figure 7(c) shows the calculated waveform of is(t) for different values of γ when Q=50. Since the rms value of ih(t) can be reduced by increasing the value of La, the degradation in the current gain can be corrected to some extent by choosing higher value of γ. Figure 8 shows the plots of current gain as a function of Q for different values of γ. A plot of the current gain predicted by (12) is also shown for direct comparison. The figure also shows the evolution of the current gain at the conventional operating point predicted by (13) for γ=1 and γ=5. It is noticed that the actual current gain at the proposed operating point is higher than that at the conventional operating point.Typical calculated waveforms ofvs(t), is1(t), ih(t), and is(t) for low-Q(Q=1;γ=1) (a) and high-Q(Q=50;γ=1) (b) operation. The effect of γ on the waveform of is(t) is illustrated by the calculated waveforms in (c) for Q=50.
(a)
(b)
(c)Figure 8
The plots of current gain as a function ofQ for different values of γ at the at the proposed operating point ωn=1 and conventional operating point ωn=ωo_c.The physical size of an inductor depends on its value, peak current, and rms current. Since, peak and rms value ofis(t) decreases with increasing La, it is intuitive to expect that the physical size of La will first reduce as La increased, reach a minimum and subsequently increase with further increase in La. To obtain a value of La resulting in minimum size, the following term is defined as its normalized size index:
(20)KLa=LaIs,pkIs,rmsLeqILeq,pkILeq,rms.
Figure 9 shows the plots of (20) for different values of Q. It is seen that for high-Q applications, La should typically be 5 times Leq.Figure 9
The plots of (20) showing the variation of KLa as a function of γ for different values of Q.


## 4. System Description

MOVPE is a highly controlled method for the deposition of semiconductor epitaxial layers and heterostructures, which are required for the development of several optoelectronic and electronic devices. The process of MOVPE involves a vapour phase reaction between metalorganic compound and a hydride gas, which are transported to a heated (around 1200°C) graphite susceptor, resulting in a growth of the desired material. A power supply is required to heat graphite susceptor (50 mm diameter and 20 mm length) to 1200°C in a quartz reactor of 80 mm diameter.


### 4.1. Work Coil and Resonant Capacitor

While recommended coil design practices suggest that the coil should be as close to the susceptor and the coil length should be larger than that of the susceptor [1], physical dimensions of the susceptor and the reactor force the inner diameter of the coil to be 100 mm. Further, since a stainless steel showerhead with nozzles for flowing gases into the reactor is attached to one end of the reactor, the maximum length of the coil is also limited to 50 mm. The coil is made up of hollow copper conductor. The number of turns, conductor diameter, and its wall thickness are optimized considering various parameters like coil resistance, power loss, electrical efficiency, required cooling water flow rate, and pressure drop. The designed coil has 4 turns of hollow copper conductor with 3/8 inch OD and SWG 19 wall thickness. The coil resistance is estimated to be 11.3 mΩ, and coil inductance is 2.70 μH. The operating frequency of 25 kHz has been fixed to ensure uniform heating of the susceptor (the skin depth of graphite is nearly 17 mm at 25 kHz). Equivalent resistance of work piece is estimated to be 16.6 mΩ. The coil is designed to carry a maximum of 1000 A rms.A bank of 6 conduction-cooled capacitors of 3μF each has been used as the resonant capacitor C. These capacitors are mounted on a cold plate, which in turn is water cooled. However, only 5 capacitors are used to get the resonant frequency of 25 kHz. The capacitor bank is kept very close to the coil to reduce the circulating path.


### 4.2. Power Circuit

A schematic circuit diagram of the power circuit is shown in Figure10, which can broadly be divided into three sections: front-end-three phase diode rectifier with filter, a dc-dc converter, and the L-LC RI. In addition to these, the actual power circuit also consists of breakers, EMI filter, and inrush-current-limiting circuit in the rectifier stage. However, these are not shown in Figure 10 for clarity.Figure 10
The power circuit and the control block diagram of a 25 kHz, 25 kW induction heating power supply using L-LC RI for MOVPE system.#The process controller, shown for the completeness, is not used in the laboratory testing of the system.The input to the power supply is 415 V, 50 Hz, and 3-phase ac. The values of filter inductor (Lf) and capacitor (Cf) in the input rectifier section are 2.2 mH and 3 mF, respectively, giving the cut-off frequency of 70 Hz.It is seen in Section2 that the L-LC RI exhibits desired behaviour only when it is operated at ωn=1. Therefore, variation of switching frequency for output power control is not permitted. Therefore, fixed-frequency control methods such as pulse width modulation (PWM) or the quantized control methods such as pulse density modulation can be used. However, the soft-switching operation of the switches in RI stage may not be guaranteed over the entire operating range, and the quantized control methods, such as PDM, result in discrete output levels and also have limited range of output power control. If, on the other hand, input voltage to the RI stage is controlled, output power control over a wider range and operational flexibility can be ascertained.  Demerits of this scheme, namely, requirement of an intermediate dc-dc converter stage and reduction of overall conversion efficiency, are sacrificed for aforementioned advantages.A dc-dc buck converter is chosen as the intermediate dc-dc converter stage. IGBTSb and diode Db constitute the main switching cell. A passive loss-less snubber circuit (composed of snubber inductor Ls, capacitors Cs1, Cs2, and diodes Ds1, Ds2, and Ds3) is used to limit switching losses in this stage. It is worthwhile to note that apart from the careful choice of the component values, described in [30], an equal attention to several practical aspects [31], such as forward recovery of snubber diodes and stray inductances, is also important for effective snubber action. Inductor Lb, capacitors Cb, Cbd, and resistor Rbd constitute a high-frequency damped low-pass filter.Two half-bridge IGBT modules SKM100GB123D are used to realize the H-bridge in the inverter section feeding square- wave voltage to the high-frequency isolation transformer. The transformerTr having 1 : 1 turns ratio has been developed using 7 pairs of EE80 core with 7 turns of the primary winding and 7 turns of the secondary winding. While the primary is wound using 0.2×50mm copper foil, oxygen-free high conductivity hollow copper conductor is used for water-cooled secondary winding. The results for Figure 9 suggest that the additional resonant inductor La should approximately be 5 times Leq for high-Q operation. Therefore, in the present system, La=13.5μH is chosen. The transformer leakage inductance is estimated to be 5 μH. The additional 8.5 μH inductance is realized by running the secondary turns around auxiliary gapped core (2 pairs of EE80), following the configuration suggested in [32].A capacitor has been placed in series with the transformer primary (not shown in Figure10) to prevent the transformer saturation in the event of unsymmetrical excitation. Table 1 summarizes the values of the components and part numbers of the semiconductor devices used in the system.Table 1
Component values and device part numbers used in the system.
Component
Value/Part number
L
f
2.2 mH
C
f
3 mF
L
b
300μH
C
b
30μF
C
b
d
150μF
R
b
d
2.2Ω
L
s
5μH
C
s
1
0.03μF
C
s
2
0.33μF
L
a
13.5μH
C
15μF
D
r
1-Dr6
VUO 82
S
b-Db
SKM100GAR123D
S
1-S4 and D1-D4
SKM100GB123D, two modules
D
s
1
,
D
s
2
,
D
s
3
DSEI 31 × 100


### 4.3. Control Circuit

The input dc voltage to the square-wave inverter is controlled to regulate the susceptor temperature. The block diagram of overall control system is also shown in Figure10. A thermocouple-type temperature sensor and process PID controller module are planned to be used for programming and controlling susceptor temperature. The output of PID module acts as the reference for the inner control loop to control the input dc voltage to the square-wave inverter by controlling the duty ratio of the buck converter. An additional phase control loop is employed to maintain the tune-in condition of the RI stage against the slow drift of resonant frequency over varied operating conditions and time. This loop senses the phase of the inverter output current using a phase detector, PD. The output of the controller drives a voltage controlled oscillator (VCO) that adjusts the switching frequency of the inverter in such a way that the inverter output current lags the voltage slightly.


## 4.1. Work Coil and Resonant Capacitor

While recommended coil design practices suggest that the coil should be as close to the susceptor and the coil length should be larger than that of the susceptor [1], physical dimensions of the susceptor and the reactor force the inner diameter of the coil to be 100 mm. Further, since a stainless steel showerhead with nozzles for flowing gases into the reactor is attached to one end of the reactor, the maximum length of the coil is also limited to 50 mm. The coil is made up of hollow copper conductor. The number of turns, conductor diameter, and its wall thickness are optimized considering various parameters like coil resistance, power loss, electrical efficiency, required cooling water flow rate, and pressure drop. The designed coil has 4 turns of hollow copper conductor with 3/8 inch OD and SWG 19 wall thickness. The coil resistance is estimated to be 11.3 mΩ, and coil inductance is 2.70 μH. The operating frequency of 25 kHz has been fixed to ensure uniform heating of the susceptor (the skin depth of graphite is nearly 17 mm at 25 kHz). Equivalent resistance of work piece is estimated to be 16.6 mΩ. The coil is designed to carry a maximum of 1000 A rms.A bank of 6 conduction-cooled capacitors of 3μF each has been used as the resonant capacitor C. These capacitors are mounted on a cold plate, which in turn is water cooled. However, only 5 capacitors are used to get the resonant frequency of 25 kHz. The capacitor bank is kept very close to the coil to reduce the circulating path.


## 4.2. Power Circuit

A schematic circuit diagram of the power circuit is shown in Figure10, which can broadly be divided into three sections: front-end-three phase diode rectifier with filter, a dc-dc converter, and the L-LC RI. In addition to these, the actual power circuit also consists of breakers, EMI filter, and inrush-current-limiting circuit in the rectifier stage. However, these are not shown in Figure 10 for clarity.Figure 10
The power circuit and the control block diagram of a 25 kHz, 25 kW induction heating power supply using L-LC RI for MOVPE system.#The process controller, shown for the completeness, is not used in the laboratory testing of the system.The input to the power supply is 415 V, 50 Hz, and 3-phase ac. The values of filter inductor (Lf) and capacitor (Cf) in the input rectifier section are 2.2 mH and 3 mF, respectively, giving the cut-off frequency of 70 Hz.It is seen in Section2 that the L-LC RI exhibits desired behaviour only when it is operated at ωn=1. Therefore, variation of switching frequency for output power control is not permitted. Therefore, fixed-frequency control methods such as pulse width modulation (PWM) or the quantized control methods such as pulse density modulation can be used. However, the soft-switching operation of the switches in RI stage may not be guaranteed over the entire operating range, and the quantized control methods, such as PDM, result in discrete output levels and also have limited range of output power control. If, on the other hand, input voltage to the RI stage is controlled, output power control over a wider range and operational flexibility can be ascertained.  Demerits of this scheme, namely, requirement of an intermediate dc-dc converter stage and reduction of overall conversion efficiency, are sacrificed for aforementioned advantages.A dc-dc buck converter is chosen as the intermediate dc-dc converter stage. IGBTSb and diode Db constitute the main switching cell. A passive loss-less snubber circuit (composed of snubber inductor Ls, capacitors Cs1, Cs2, and diodes Ds1, Ds2, and Ds3) is used to limit switching losses in this stage. It is worthwhile to note that apart from the careful choice of the component values, described in [30], an equal attention to several practical aspects [31], such as forward recovery of snubber diodes and stray inductances, is also important for effective snubber action. Inductor Lb, capacitors Cb, Cbd, and resistor Rbd constitute a high-frequency damped low-pass filter.Two half-bridge IGBT modules SKM100GB123D are used to realize the H-bridge in the inverter section feeding square- wave voltage to the high-frequency isolation transformer. The transformerTr having 1 : 1 turns ratio has been developed using 7 pairs of EE80 core with 7 turns of the primary winding and 7 turns of the secondary winding. While the primary is wound using 0.2×50mm copper foil, oxygen-free high conductivity hollow copper conductor is used for water-cooled secondary winding. The results for Figure 9 suggest that the additional resonant inductor La should approximately be 5 times Leq for high-Q operation. Therefore, in the present system, La=13.5μH is chosen. The transformer leakage inductance is estimated to be 5 μH. The additional 8.5 μH inductance is realized by running the secondary turns around auxiliary gapped core (2 pairs of EE80), following the configuration suggested in [32].A capacitor has been placed in series with the transformer primary (not shown in Figure10) to prevent the transformer saturation in the event of unsymmetrical excitation. Table 1 summarizes the values of the components and part numbers of the semiconductor devices used in the system.Table 1
Component values and device part numbers used in the system.
Component
Value/Part number
L
f
2.2 mH
C
f
3 mF
L
b
300μH
C
b
30μF
C
b
d
150μF
R
b
d
2.2Ω
L
s
5μH
C
s
1
0.03μF
C
s
2
0.33μF
L
a
13.5μH
C
15μF
D
r
1-Dr6
VUO 82
S
b-Db
SKM100GAR123D
S
1-S4 and D1-D4
SKM100GB123D, two modules
D
s
1
,
D
s
2
,
D
s
3
DSEI 31 × 100


## 4.3. Control Circuit

The input dc voltage to the square-wave inverter is controlled to regulate the susceptor temperature. The block diagram of overall control system is also shown in Figure10. A thermocouple-type temperature sensor and process PID controller module are planned to be used for programming and controlling susceptor temperature. The output of PID module acts as the reference for the inner control loop to control the input dc voltage to the square-wave inverter by controlling the duty ratio of the buck converter. An additional phase control loop is employed to maintain the tune-in condition of the RI stage against the slow drift of resonant frequency over varied operating conditions and time. This loop senses the phase of the inverter output current using a phase detector, PD. The output of the controller drives a voltage controlled oscillator (VCO) that adjusts the switching frequency of the inverter in such a way that the inverter output current lags the voltage slightly.


## 5. Results

The photograph of the developed induction heating power supply, being tested in laboratory to heat a graphite susceptor, is shown in Figure11. The graphite susceptor heated to 1200°C in air is shown in the inset.Figure 11
The photograph of the developed induction heating power supply. The graphite susceptor heated to 1200°C is shown in the inset.The inverter output voltage,vs, (trace 1, 200 V/div) and current waveforms, is, (trace 2, 25 A/div) are shown in Figure 12(a). The nature of current waveform matches with the predicted waveform given in Figure 7. The waveforms of collector-to-emitter (trace 1, 200 V/div) and gate-to-emitter (trace 2, 10 V/div) voltage of an IGBT in H-bridge inverter during turn-on and turn-off transitions are shown in Figures 12(b) and 12(c), respectively, demonstrating the soft-switching. Figure 12(d) shows the waveforms of vs (trace 1, 200 V/div) and the voltage across the work coil (trace 2, 200 V/div) showing that only fundamental component of input square-wave voltage has been passed to the work coil.Experimental waveforms. (a)vs (trace 1, 200 V/div) and is (trace 2, 25 A/div); x-scale: 5 μs/div. (b) Waveforms at turn-on: IGBT collector-to-emitter voltage (trace 1, 200 V/div) and gate-to-emitter voltage (trace 2, 10 V/div) x-scale: 500 ns/div. (c) Waveforms at turn-off: IGBT collector-to-emitter voltage (trace 1, 200 V/div) and gate-to-emitter voltage (trace 2, 10 V/div) x-scale: 500 ns/div. (d) vs (trace 1, 200 V/div) and the voltage across the work coil (trace 2, 200 V/div) x-scale: 10 μs/div.
(a)
(b)
(c)
(d)


## 6. Conclusion

The paper reports the various issues in the development of an IH power supply for MOVPE application using L-LC RI. The fundamental frequency ac analysis of L-LC RI is revisited, and it is shown that the converter exhibits enhanced current gain and near-unity power factor operation when operated at the resonant frequency. Further analysis of the circuit with square-wave voltage source highlights the effect of the auxiliary inductor on the source current waveform leading to the optimum choice for the value of auxiliary inductor. The requirements of MOVPE system demanding a 25 kW, 25 kHz IH power supply to heat a graphite susceptor to 1200°C, the configuration of developed IH system, and experimental results are presented, thereby demonstrating the suitability of L-LC RI operating at its resonant frequency for this application.



---

*Source: 584129-2013-07-17.xml*