# A Low Power ROIC with Extended Counting ADC Based on Circuit Noise Analysis for Sensor Arrays in IoT System

Authors: Ye Zhou; Wengao Lu; Shanzhe Yu; Dunshan Yu; Yacong Zhang; Zhongjian Chen
Journal: Journal of Sensors
 (2022)

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## Abstract

As the Internet of Things (IoT) is rapidly integrated into our daily life, the demand for high performance readout integrated circuit (ROIC) design for sensor arrays is boosting. This paper presents a low power, low noise ROIC with 14-bit column-parallel extended counting (EC) ADCs for sensor arrays targeting the IoT applications. The proposed EC-ADC adopts a pseudodifferential architecture to cancel even-order nonlinearity. The analog front-end is aGm stage, which employs a current-reuse topology to boost the transconductance and reduce noise without increasing current consumption. The upper 9-bit conversion is implemented during integration, and the residual voltage is converted by a 5-bit single-slope (SS) ADC, where the comparator is reused. A ping-pong integrator is proposed to reduce the reset time and improve linearity, eliminating the power-hungry CTIA structure. The ROIC is designed in 0.18 μm 1P5M CMOS process for a 640×480 sensor array. Power consumption of the ROIC is 33 mW, and each column ADC consumes 40.1 μW. Simulation results show an input-referred noise of 0.89 LSB (1.74 μVrms), an integral nonlinearity of +0.92/-0.70 LSB, an ENOB of 12.87 bits, and a FoM of 131.1 fJ/step.


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## Body

## 1. Introduction

The growing interest in the field of smart sensors has set new limits on realizing different sensors that are capable to capture myriad types of signals. These applications are often size constrained, while requiring high energy efficiency and stability [1–4]. With the advantages of low cost, small size, and good uniformity, uncooled infrared focal plane arrays (IRFPA) can be applied in IoT applications, such as composition analysis, smart monitoring, and spectral analysis [5–12].Currently, the mainstream IRFPAs adopt vanadium oxide (VOX) [5, 6] or amorphous silicon (α-Si) microbolometers [7, 8], or silicon diodes as the thermosensitive sensors [9, 10]. Compared with VOX/α-Si microbolometers, silicon diodes are fabricated with monocrystalline silicon and have the advantages of CMOS process compatibility, low noise, high uniformity, and huge potential for pixel size reduction [11]. However, the temperature sensitivity of silicon diodes is lower than that of VOX/α-Si microbolometers, which leads to smaller signal voltage [12]. Therefore, the noise of the readout integrated circuit (ROIC) will have more severe impact and need to be reduced to achieve high SNR. Increasing power consumption is a common way to lower noise. But the sensors of the IRFPA are sensitive to substrate temperature, which means decreasing power consumption of the ROIC is important to minimize the temperature drift. Besides, the miniaturized and portable development tendency also poses strict requirements on power dissipation. In addition, the ROIC needs to attain reasonable linearity to reduce fixed background noise [13]. In summary, the ROIC for silicon diode IRFPA needs to achieve low noise and good linearity under high power efficiency.To meet these requirements, the integrated analog-to-digital converter (ADC) plays a key role. In recent years, the extended counting (EC) ADC has been developed as an excellent candidate among various kinds of the ADC solutions for large-scale sensor array ROICs [14–18]. Compared with other ADC types such as successive approximate register (SAR) ADC, delta-sigma (Δ-Σ) ADC, and SS-ADC, EC-ADC achieves a good balance between bit-depth, conversion speed, and area. EC-ADCs implement coarse A/D conversion through multiple reset operation with a 1/1.5-bit feedback digital-to-analog converter (DAC) during the folding integration (FI) period. And the analog residue is then quantified during the EC period. This combination can maximize the integration time, thereby reducing noise while enhancing bit-depth with better power efficiency. However, to improve linearity and provide source-and-sink integration current during the integration period, a capacitive transimpedance amplifier (CTIA) is often needed in the integrator. The CTIA block is power-hungry to achieve high slew rate and bandwidth. In [15], a folded cascode operational amplifier (op-amp) with gain boosting is adopted to improve the linearity of the CTIA, while the power consumption is too high for large-scale sensor array application. In [16], a current compensation technique is proposed to provide compensation current during the FI period, thus easing the power budget. But an extra high frequency clock is introduced which will generate jitter noise and increase design complexity and digital crosstalk. Some CTIA-less architectures have also been reported. In [17], a passive integrator which included a capacitor and an injection transistor is used to replace the power-hungry CTIA structure. Yet the input signal range is unidirectional, which impairs input signal swing and signal-to-noise ratio (SNR). Moreover, there is reset time caused by the reset operation during integration, which will further degrade the accuracy and linearity [18].In this paper, a low power ROIC with 14-bit column-wise EC-ADCs targeting a640×480 silicon diode uncooled IRFPA under 50 Hz frame rate is proposed. A FI ADC and a single-slope (SS) ADC are combined for the upper 9-bit and lower 5-bit conversions, respectively. As the analog front-end, the Gm stage is the main power consumption and noise source. We adopt a pseudodifferential current-reuse structure to boost the transconductance and lower noise without increasing current consumption. A passive ping-pong integrator architecture comprising two switched capacitors with buffered direct injection (BDI) structure is proposed to replace the power-hungry CTIA, saving energy while keeping high linearity. The comparator with positive feedback (PFB) technique is shared between the FI and the SS-ADC to reduce power dissipation and circuit complexity. The proposed ROIC achieves a good balance between power efficiency, resolution, and noise performance. The remaining of this paper is organized as follows. The ROIC system architecture including system overview, circuit diagram, and its operation principle is described in Section 2. The detailed circuit design, noise, and linearity analysis are presented in Section 3. Chip layout and postsimulation results are given in Section 4. Section 5 concludes this paper.


## 2. Proposed EC-ADC-Based ROIC Architecture

### 2.1. ROIC Structure Overview

Figure1 shows a conceptual block diagram of the proposed ROIC. Each column-wise ADC consists of a 10-bit (including one redundant bit) FI ADC and a 5-bit SS-ADC. In step 1, the FI ADC converts the differential voltage into current and implements coarse A/D conversion through charging-and-resetting process during the FI period. The reset or folding times are proportional to the integrating current and recorded in the counter. In step 2, integration stops, VTH starts to ramp down, and the residue voltage is further quantified by the SS-ADC. The upper 10-bit and lower 5-bit are then sent out serially to generate the final conversion results.Figure 1
Conceptual block diagram of the proposed ROIC.The circuit diagram of the proposed EC-ADC-based ROIC and its signal flow are shown in Figure2. This circuit is a pseudodifferential implementation of Figure 1. In this scheme, six silicon diodes biased at constant current are connected in series as the thermosensitive sensors, and the diode voltages change with temperature. When radiation arrives, the temperature of the sensitive pixel changes while the temperature of the blind pixel is unaffected. As a result, the differential voltages are measured and quantified row by row. To minimize the spatial nonuniformity of the diodes, on-chip calibration (OCC) is adopted to provide calibrated common-mode voltage VCM to the Gm stage. The OCC block is a low noise 6-bit DAC, where the noise is less than 1 μVrms (0.5 LSB). The proposed ROIC adopts pseudodifferential topology, which can reduce the relative quantization step size, cancel common-mode disturbances such as substrate temperature variation, and improve linearity by subtracting even-order distortion. Each column readout circuit includes a Gm stage and a differential EC-ADC. The EC-ADC uses ping-pong integrator instead of CTIA structure to reduce power dissipation while maintaining good linearity. A BDI structure, which is composed of a PMOS transistor and a feedback amplifier, is employed to offer higher current injection efficiency and stabilize the voltage at the output of Gm stage.Figure 2
Circuit diagram of the proposed ROIC and its signal flow in one column.Figure3 shows the timing diagram of the proposed ROIC. In the i-th row period, after autozero operation, the voltage difference between the sensing and blind pixels is converted into differential integration currents. VINTPN stars to ramp up during integration, in which the integrator adopted ping-pong structure. When one capacitor is connected to the comparator for integration, the other is reset to VL. Each time VINTPN reaches the threshold voltage VH, the comparators are triggered to switch S13 and S24. Such switching between integration and reset can reduce reset time and improve linearity, which will be further explained in Section 3.3. When ΦINT and ΦMSB are turned off, integration stops, and there is a residue voltage VresPN. Then, ΦLSB is turned on, VTH starts to ramp down from VH to VL, and the comparator flips when VTH reaches VresPN. Note that ΦINT, ΦMSB, and ΦLSB have different operation time to avoid false trigger and reduce the effects of charge injection. Finally, the MSBs and LSBs are combined and read out serially through the registers.Figure 3
Timing diagram of the proposed ROIC.In this design, a differentialGm stage is directly connected to diodes to provide high input impedance to sense the voltage signals, which can eliminate preamplifier for better hardware efficiency. The FI ADC has one redundant bit and performs conversion while integration, which can increase the integration time to achieve better noise performance compared with conversion-after-integration method. The ramp voltage for fine conversion is generated by a global ramp generator which is shared by the whole arrays. The comparator is also shared between the FI and the SS-ADC to further save energy and decrease system complexity.


### 2.2. Circuit Noise Analysis

This section focuses on noise analysis and optimization of the column readout circuit. Since the column ADC is directly connected to the diode sensors, it is most convenient to calculate the total noise as input-referred noise. To minimize the influence of the circuit noise on the whole image system, it is essential to reduce the circuit noise to a low level, for example, less than 1 LSB root-mean-square (RMS) noise under 14-bit resolution. The circuit noise contribution comes from theGm stage (including thermal and flicker noise), the reset (kTC) noise and the comparator (VCMP2¯), and need to be integrated over a bandwidth of 1/2TINT [19]. The input-referred noise contributed by Gm can be expressed as
(1)Vn,Gm2¯=∫f11/2TINT8kT3Gm+KCoxWGmLGm1fdf,where f1 is chosen to avoid divergence at f=0 and is equal to 1/4ntframe, where n is the number of frames being sampled [20]. The noise charge accumulated on the integrators from kTC and comparator is NKTCU and VCMP2¯CU2/2TINT, respectively, where N is the number of triggers during the folding integration period (TINT). And this noise charge divided by TINT equals to noise current. Then, the input-referred noise from kTC and comparator is given by
(2)Vn,kTC,CMP2¯=1Gm2NkTCU+VCMP2CU212TINT1TINT2=1Gm2TINT2/CU2NkTCU+VCMP2¯12TINT,where k is Boltzmann’s constant and T is the circuit absolute temperature. In this design, TINT/CU is in the order of tens of megohm, and Gm is with the order of hundreds of siemens. In general, the Gm2TINT2/CU2 term is rather large, which means (2) is much smaller than (1) and the Gm stage is the main noise source of the circuit. According to (1) and (2), increasing Gm is of great importance for decreasing the overall circuit noise.


## 2.1. ROIC Structure Overview

Figure1 shows a conceptual block diagram of the proposed ROIC. Each column-wise ADC consists of a 10-bit (including one redundant bit) FI ADC and a 5-bit SS-ADC. In step 1, the FI ADC converts the differential voltage into current and implements coarse A/D conversion through charging-and-resetting process during the FI period. The reset or folding times are proportional to the integrating current and recorded in the counter. In step 2, integration stops, VTH starts to ramp down, and the residue voltage is further quantified by the SS-ADC. The upper 10-bit and lower 5-bit are then sent out serially to generate the final conversion results.Figure 1
Conceptual block diagram of the proposed ROIC.The circuit diagram of the proposed EC-ADC-based ROIC and its signal flow are shown in Figure2. This circuit is a pseudodifferential implementation of Figure 1. In this scheme, six silicon diodes biased at constant current are connected in series as the thermosensitive sensors, and the diode voltages change with temperature. When radiation arrives, the temperature of the sensitive pixel changes while the temperature of the blind pixel is unaffected. As a result, the differential voltages are measured and quantified row by row. To minimize the spatial nonuniformity of the diodes, on-chip calibration (OCC) is adopted to provide calibrated common-mode voltage VCM to the Gm stage. The OCC block is a low noise 6-bit DAC, where the noise is less than 1 μVrms (0.5 LSB). The proposed ROIC adopts pseudodifferential topology, which can reduce the relative quantization step size, cancel common-mode disturbances such as substrate temperature variation, and improve linearity by subtracting even-order distortion. Each column readout circuit includes a Gm stage and a differential EC-ADC. The EC-ADC uses ping-pong integrator instead of CTIA structure to reduce power dissipation while maintaining good linearity. A BDI structure, which is composed of a PMOS transistor and a feedback amplifier, is employed to offer higher current injection efficiency and stabilize the voltage at the output of Gm stage.Figure 2
Circuit diagram of the proposed ROIC and its signal flow in one column.Figure3 shows the timing diagram of the proposed ROIC. In the i-th row period, after autozero operation, the voltage difference between the sensing and blind pixels is converted into differential integration currents. VINTPN stars to ramp up during integration, in which the integrator adopted ping-pong structure. When one capacitor is connected to the comparator for integration, the other is reset to VL. Each time VINTPN reaches the threshold voltage VH, the comparators are triggered to switch S13 and S24. Such switching between integration and reset can reduce reset time and improve linearity, which will be further explained in Section 3.3. When ΦINT and ΦMSB are turned off, integration stops, and there is a residue voltage VresPN. Then, ΦLSB is turned on, VTH starts to ramp down from VH to VL, and the comparator flips when VTH reaches VresPN. Note that ΦINT, ΦMSB, and ΦLSB have different operation time to avoid false trigger and reduce the effects of charge injection. Finally, the MSBs and LSBs are combined and read out serially through the registers.Figure 3
Timing diagram of the proposed ROIC.In this design, a differentialGm stage is directly connected to diodes to provide high input impedance to sense the voltage signals, which can eliminate preamplifier for better hardware efficiency. The FI ADC has one redundant bit and performs conversion while integration, which can increase the integration time to achieve better noise performance compared with conversion-after-integration method. The ramp voltage for fine conversion is generated by a global ramp generator which is shared by the whole arrays. The comparator is also shared between the FI and the SS-ADC to further save energy and decrease system complexity.


## 2.2. Circuit Noise Analysis

This section focuses on noise analysis and optimization of the column readout circuit. Since the column ADC is directly connected to the diode sensors, it is most convenient to calculate the total noise as input-referred noise. To minimize the influence of the circuit noise on the whole image system, it is essential to reduce the circuit noise to a low level, for example, less than 1 LSB root-mean-square (RMS) noise under 14-bit resolution. The circuit noise contribution comes from theGm stage (including thermal and flicker noise), the reset (kTC) noise and the comparator (VCMP2¯), and need to be integrated over a bandwidth of 1/2TINT [19]. The input-referred noise contributed by Gm can be expressed as
(1)Vn,Gm2¯=∫f11/2TINT8kT3Gm+KCoxWGmLGm1fdf,where f1 is chosen to avoid divergence at f=0 and is equal to 1/4ntframe, where n is the number of frames being sampled [20]. The noise charge accumulated on the integrators from kTC and comparator is NKTCU and VCMP2¯CU2/2TINT, respectively, where N is the number of triggers during the folding integration period (TINT). And this noise charge divided by TINT equals to noise current. Then, the input-referred noise from kTC and comparator is given by
(2)Vn,kTC,CMP2¯=1Gm2NkTCU+VCMP2CU212TINT1TINT2=1Gm2TINT2/CU2NkTCU+VCMP2¯12TINT,where k is Boltzmann’s constant and T is the circuit absolute temperature. In this design, TINT/CU is in the order of tens of megohm, and Gm is with the order of hundreds of siemens. In general, the Gm2TINT2/CU2 term is rather large, which means (2) is much smaller than (1) and the Gm stage is the main noise source of the circuit. According to (1) and (2), increasing Gm is of great importance for decreasing the overall circuit noise.


## 3. Circuit Design

### 3.1. PseudodifferentialGm Stage

To achieve higher noise efficiency of theGm stage, a current-reuse topology is adopted, which is also known as the inverter-based structure, as shown in Figure 4. PMOS and NMOS input-pairs are stacked to double the transconductance without increasing power consumption. Instead of adding extra current sources at the output ports as shown in Figure 5, the proposed design incorporates the biasing source into the Gm stage, which further increases the Gm. Therefore, the thermal noise of the Gm stage is reduced and the noise in (2) is also suppressed. The area of input-pairs is doubled compared with traditional structure, which further restrain the flicker noise according to (1). The source resistors RS are utilized to improve linearity through negative feedback.Figure 4
Proposed circuit schematic of the current-reuseGm stage, incorporating the outside current source to increase the transconductance.Figure 5
TraditionalGm stage with extra biasing current source.In Figure4, IBP=20 μA, IBN=15 μA, and IINT0=2.5 μA. The PMOS and NMOS transistors of the input pairs are sized 2.56 mm/0.5 μm and 2.56 mm/1 μm, respectively. The resistance of RS is 4 kΩ. The integration current (IINTPN) is the difference between IPRL and INRL. The direction of IINTPN is continuously flowing out during integration. In this pseudodifferential design, for a 16 mVP (32 mVPP) signal of VSIG, IINTPN has a swing of ±1.75 μA under 2.5 μA common-mode current, which is translated into ±70% frequency swing range by the following EC-ADC.


### 3.2. Ping-Pong Integrator and Nonlinearity Analysis

During the coarse conversion period, the integration current is converted into frequency signal and recorded in the counter. Each time the nodeVINTPN reaches the threshold voltage VH, the comparator will activate, and VINTPN is reset to VL. In this charging-and-resetting process, the amount of charge QINT accumulated on the integration capacitor during one conversion period is expressed as below:
(3)QINT=IINTTINT=N×CUVR+IINTtRST+CUVres,where N is the number of triggers, tRST is reset time, and VR is equal to (VH−VL). Ideally, the charge packet CUVR is used to quantify the signal charge QINT, in which the correlation between signal charge and counting number is written as QINT=NidealCUVR, where Nideal is the ideal resetting times if there is no reset time and residual voltage. However, as shown in Figure 5(b), the existence of tRST and Vres causes nonlinearity. During the FI period, the nonlinearity of quantization number ΔN can be calculated as follows:
(4)N=IINTTINT−CUVεCUVR+IINTtRST≈IINTTINTCUVR+IINTtRST,(5)∆N=Nideal−N=IINTTINTCUVR−IINTTINTCUVR+IINTtRST=IINTTINTCUVR×11+CUVR/IINTtRST.An approximation is made in (4) because IINTTINT is normally much larger than CUVres and Vres is further processed by the SS-ADC in this work. For a 50 Hz 640×480 array, TINT is set as 40 μs. The charge packet CUVR is decided by the input range and ADC resolution and is set to be about 0.52 pC in this design. According to (5), ΔN is positively related to IINT and tRST, which means that the nonlinearity will increase with integration current and reset time.In conventional reset operation as shown in Figure6(a), the reset time tRST is composed of three parts: the time between when VINT reaches VH and turning on the reset switch (td1), charging time of CU (tCH), and the time between the end of the charging and turning off the reset switch (td2). VINT will not start rising until another period (tCH+td2) passed, and this period increases with the increasing of signal current [18]. Decreasing the propagation delay of the comparator can shorten the reset time, but this requires higher power consumption. Moreover, in conventional reset operation, td1 and tCH are necessary periods allowing VINT to reset fully and must maintain a reasonable length.Figure 6
Illustration of the timing diagram of (a) the conventional reset operation and (b) the proposed ping-pong reset operation. Comparison of (c) the reset time and (d) INL performance between proposed ping-pong structure and conventional structure under the same power consumption.
(a)(b)(c)(d)To solve the above problems, a ping-pong architecture is adopted to reduce the reset time without increasing power consumption. Different from conventional structure,S13 and S24 will turn on alternatively when VINTPN reaches the threshold voltage VH. Because reset operation of the capacitor is completed before connecting to the comparator, charging time tCH and the restart delay td2 due to turning off the reset switch in conventional structure is eliminated, as shown in Figure 6(b). The time for the node VINTPN to stabilize (tstb) is less than 200 ps, which is much smaller than tCH+td2 in conventional structure. Moreover, as there is not charging time limitation in the proposed ping-pong architecture, td1 can also be further decreased. As a result, the total reset time tRST is much reduced. Figure 6(c) shows the comparison of tRST between the proposed and conventional structure in the case of the same power dissipation. It can be observed that tRST is much reduced with the proposed ping-pong operation, leading to higher linearity. Figure 6(d) shows linearity comparison. The integral nonlinearity (INL) is reduced from +2.45/-4.75 LSB to +0.92/-0.70 LSB, which further proves the effectiveness of this design.The integration capacitorCU is implemented with 645-fF MIM capacitor, which is sized as 2×4 μm/30 μm. The switches S1~S4 are sized to minimum values to minimize unwanted charge injection. Using a CTIA structure can also achieve high linearity, but the power consumption will increase to tens of μW because CTIA needs to provide integration current. In this design, each feedback amplifier consumes only 100 nA. In general, the proposed ping-pong integrator greatly reduces power consumption while keeping high linearity.Although the mismatch between the two capacitors in the proposed ping-pong integrator may introduce extra nonlinearity, this nonlinearity is quite limited. When considering mismatch, the capacitance isCu1 and Cu2, respectively. And the injected charge of every two reset operations is (CU1VR+CU2VR), which is independent of the mismatch. Only the last reset operation introduces nonlinearity, and the three sigma of capacitance mismatch is 0.5% according to the statistical model provided by foundry. This 0.5% mismatch leads to 0.5%×32=0.16 LSB nonlinearity in the proposed upper 9-bit and lower 5-bit EC-ADC. Monte-Carlo simulation results are also given in Section 4 for verification.


### 3.3. Comparator with PFB Technique

To reduce the propagation delay (td1) during the ping-pong reset operation, the comparator is required to response quickly when VINT reaches the threshold voltage. Figure 7 illustrates the proposed comparator with positive feedback (PFB) circuit. When ΦRST drops down and integration starts, VP is lower than VN, VFB is charged to VDD, and M5 is turned on while M6 and M3 are turned off, which diminish the leakage current of the second stage. When VP rises and reaches VN, M3 is turned on, and the race between M3 and M6 is avoided; thus, VFB drops faster. The feedback path of M4 and the AND gate also accelerates dropping of VFB. As a result, the delay of the comparator is reduced. When VFB drops to zero, Vout turns high, and VP is reset to VL by the ping-pong integrator. Note that when VFB is low, M6 will turn on through the delay path (after VP is reset to VL), and VFB will be pulled back to VDD. After precharging VFB, M6 will be turned off again through the delay path, and an operating cycle is completed. The proposed PFB circuit can reduce the delay and the static power of the comparator. In this design, the ground of the comparators is separated into VSSA2 and VSSD to reduce the ground crosstalk. As shown in Figure 8, the maximum current consumption is less than 1.14 μA. And the comparator consumes less power when the number of triggers is lower.Figure 7
Comparator structure with positive feedback circuit.Figure 8
Variation of the comparator current of different number of triggers.


## 3.1. PseudodifferentialGm Stage

To achieve higher noise efficiency of theGm stage, a current-reuse topology is adopted, which is also known as the inverter-based structure, as shown in Figure 4. PMOS and NMOS input-pairs are stacked to double the transconductance without increasing power consumption. Instead of adding extra current sources at the output ports as shown in Figure 5, the proposed design incorporates the biasing source into the Gm stage, which further increases the Gm. Therefore, the thermal noise of the Gm stage is reduced and the noise in (2) is also suppressed. The area of input-pairs is doubled compared with traditional structure, which further restrain the flicker noise according to (1). The source resistors RS are utilized to improve linearity through negative feedback.Figure 4
Proposed circuit schematic of the current-reuseGm stage, incorporating the outside current source to increase the transconductance.Figure 5
TraditionalGm stage with extra biasing current source.In Figure4, IBP=20 μA, IBN=15 μA, and IINT0=2.5 μA. The PMOS and NMOS transistors of the input pairs are sized 2.56 mm/0.5 μm and 2.56 mm/1 μm, respectively. The resistance of RS is 4 kΩ. The integration current (IINTPN) is the difference between IPRL and INRL. The direction of IINTPN is continuously flowing out during integration. In this pseudodifferential design, for a 16 mVP (32 mVPP) signal of VSIG, IINTPN has a swing of ±1.75 μA under 2.5 μA common-mode current, which is translated into ±70% frequency swing range by the following EC-ADC.


## 3.2. Ping-Pong Integrator and Nonlinearity Analysis

During the coarse conversion period, the integration current is converted into frequency signal and recorded in the counter. Each time the nodeVINTPN reaches the threshold voltage VH, the comparator will activate, and VINTPN is reset to VL. In this charging-and-resetting process, the amount of charge QINT accumulated on the integration capacitor during one conversion period is expressed as below:
(3)QINT=IINTTINT=N×CUVR+IINTtRST+CUVres,where N is the number of triggers, tRST is reset time, and VR is equal to (VH−VL). Ideally, the charge packet CUVR is used to quantify the signal charge QINT, in which the correlation between signal charge and counting number is written as QINT=NidealCUVR, where Nideal is the ideal resetting times if there is no reset time and residual voltage. However, as shown in Figure 5(b), the existence of tRST and Vres causes nonlinearity. During the FI period, the nonlinearity of quantization number ΔN can be calculated as follows:
(4)N=IINTTINT−CUVεCUVR+IINTtRST≈IINTTINTCUVR+IINTtRST,(5)∆N=Nideal−N=IINTTINTCUVR−IINTTINTCUVR+IINTtRST=IINTTINTCUVR×11+CUVR/IINTtRST.An approximation is made in (4) because IINTTINT is normally much larger than CUVres and Vres is further processed by the SS-ADC in this work. For a 50 Hz 640×480 array, TINT is set as 40 μs. The charge packet CUVR is decided by the input range and ADC resolution and is set to be about 0.52 pC in this design. According to (5), ΔN is positively related to IINT and tRST, which means that the nonlinearity will increase with integration current and reset time.In conventional reset operation as shown in Figure6(a), the reset time tRST is composed of three parts: the time between when VINT reaches VH and turning on the reset switch (td1), charging time of CU (tCH), and the time between the end of the charging and turning off the reset switch (td2). VINT will not start rising until another period (tCH+td2) passed, and this period increases with the increasing of signal current [18]. Decreasing the propagation delay of the comparator can shorten the reset time, but this requires higher power consumption. Moreover, in conventional reset operation, td1 and tCH are necessary periods allowing VINT to reset fully and must maintain a reasonable length.Figure 6
Illustration of the timing diagram of (a) the conventional reset operation and (b) the proposed ping-pong reset operation. Comparison of (c) the reset time and (d) INL performance between proposed ping-pong structure and conventional structure under the same power consumption.
(a)(b)(c)(d)To solve the above problems, a ping-pong architecture is adopted to reduce the reset time without increasing power consumption. Different from conventional structure,S13 and S24 will turn on alternatively when VINTPN reaches the threshold voltage VH. Because reset operation of the capacitor is completed before connecting to the comparator, charging time tCH and the restart delay td2 due to turning off the reset switch in conventional structure is eliminated, as shown in Figure 6(b). The time for the node VINTPN to stabilize (tstb) is less than 200 ps, which is much smaller than tCH+td2 in conventional structure. Moreover, as there is not charging time limitation in the proposed ping-pong architecture, td1 can also be further decreased. As a result, the total reset time tRST is much reduced. Figure 6(c) shows the comparison of tRST between the proposed and conventional structure in the case of the same power dissipation. It can be observed that tRST is much reduced with the proposed ping-pong operation, leading to higher linearity. Figure 6(d) shows linearity comparison. The integral nonlinearity (INL) is reduced from +2.45/-4.75 LSB to +0.92/-0.70 LSB, which further proves the effectiveness of this design.The integration capacitorCU is implemented with 645-fF MIM capacitor, which is sized as 2×4 μm/30 μm. The switches S1~S4 are sized to minimum values to minimize unwanted charge injection. Using a CTIA structure can also achieve high linearity, but the power consumption will increase to tens of μW because CTIA needs to provide integration current. In this design, each feedback amplifier consumes only 100 nA. In general, the proposed ping-pong integrator greatly reduces power consumption while keeping high linearity.Although the mismatch between the two capacitors in the proposed ping-pong integrator may introduce extra nonlinearity, this nonlinearity is quite limited. When considering mismatch, the capacitance isCu1 and Cu2, respectively. And the injected charge of every two reset operations is (CU1VR+CU2VR), which is independent of the mismatch. Only the last reset operation introduces nonlinearity, and the three sigma of capacitance mismatch is 0.5% according to the statistical model provided by foundry. This 0.5% mismatch leads to 0.5%×32=0.16 LSB nonlinearity in the proposed upper 9-bit and lower 5-bit EC-ADC. Monte-Carlo simulation results are also given in Section 4 for verification.


## 3.3. Comparator with PFB Technique

To reduce the propagation delay (td1) during the ping-pong reset operation, the comparator is required to response quickly when VINT reaches the threshold voltage. Figure 7 illustrates the proposed comparator with positive feedback (PFB) circuit. When ΦRST drops down and integration starts, VP is lower than VN, VFB is charged to VDD, and M5 is turned on while M6 and M3 are turned off, which diminish the leakage current of the second stage. When VP rises and reaches VN, M3 is turned on, and the race between M3 and M6 is avoided; thus, VFB drops faster. The feedback path of M4 and the AND gate also accelerates dropping of VFB. As a result, the delay of the comparator is reduced. When VFB drops to zero, Vout turns high, and VP is reset to VL by the ping-pong integrator. Note that when VFB is low, M6 will turn on through the delay path (after VP is reset to VL), and VFB will be pulled back to VDD. After precharging VFB, M6 will be turned off again through the delay path, and an operating cycle is completed. The proposed PFB circuit can reduce the delay and the static power of the comparator. In this design, the ground of the comparators is separated into VSSA2 and VSSD to reduce the ground crosstalk. As shown in Figure 8, the maximum current consumption is less than 1.14 μA. And the comparator consumes less power when the number of triggers is lower.Figure 7
Comparator structure with positive feedback circuit.Figure 8
Variation of the comparator current of different number of triggers.


## 4. Postlayout Simulation Results

Figures9 and 10 show the block diagram and layout of the 640×480 silicon diode uncooled IRFPA chip, respectively. The proposed ROIC is designed in 0.18 μm 1P5M CMOS process. The chip size is 1.95mm×1.50mm. The blind pixels and the ramp generator are placed on the left side while the row selector and timing control circuit being at the right side to reduce the digital crosstalk. Part of the input-pairs (not labeled) of Gm stage has been placed in the pixels, which shortens the analog path. Figure 11 shows the power consumption breakdown, total power consumption of the proposed ROIC is 33 mW, and each column ADC consumes 40.1 μW (including Gm and EC-ADC). As the front end, Gm takes up a significant portion of the overall power budget to reduce the total noise. The power consumption of the Gm stage can be reduced at the cost of degraded noise and linearity performance. Figure 12 shows the FFT spectrum with a 4.639 kHz 32 mVPP sinusoid input signal under 25 kS/s sampling rate. The SFDR, SNR, SNDR, and ENOB are 88.56 dB, 84.03 dB, 79.21 dB, and 12.87 bits, respectively. Figure 13 shows the Monte-Carlo analysis of ENOB. The minimum and average ENOB are 12.61 bits and 12.74 bits. INL of the proposed ROIC in 15-run Monte-Carlo simulation is presented in Figure 14. All curves are within +1.49/-1.41 LSB, and the typical INL is +0.92/-0.70 LSB. The average and standard deviation of maximum |INL| is 1.23/0.22 LSB, which indicates the impact of mismatch is limited. Figure 15 shows the input-referred RMS noise under different ADC output code; each point is calculated as the standard deviation of 128 samples taken in transient noise simulation. The reset (kTC) noise increases with number of triggers, while the main noise source is the Gm stage. Benefited from low noise design of the Gm stage, the RMS noise is kept within 1.74 μV (0.89 LSB).Figure 9
Block diagram of the chip core.Figure 10
Chip layout.Figure 11
Power consumption breakdown average to per column.Figure 12
FFT spectrum in postsimulation.Figure 13
The ENOB performance in Monte-Carlo postsimulation.Figure 14
The INL performance in Monte-Carlo postsimulation.Figure 15
Input-referred noise performance in postsimulation.The performance of the proposed ROIC is summarized in Table1. The total power consumption is 33 mW, including the column ADC and ADC peripheral circuit. Table 2 presents the performance comparison with previous works. Various figures of merits (FoM) are calculated for a fair performance comparison in circuit energy efficiency. FoM1 and FoM2 are used for evaluating the performance of the ROIC, and FoM3 is used for representing the performance of column ADC. For various FoMs, this work shows state-of-art level performance: FoM1 of 131.1 fJ/step, FoM2 of 2.15 nJ, and FoM3 of 214.3 fJ/step. Regarding FoM1, FoM3, INL, and ENOB, this work shows competitive performance compared to other works. It can be seen that the proposed ROIC is more power efficient while maintaining high linearity and noise performance.Table 1
ROIC performance summary.
ParameterValueProcess0.18μmPower supply1.8 VArray size640×480Frame rate50 HzPixel rate15.36 Mp/sSensor materialSilicon diodesPixel pitch25μmROIC input range±16 mVADC resolution14 bitsINL+0.92/-0.70 LSBInput-referred RMS noise1.74μVrms/0.89 LSBROIC power consumption33 mWPower consumption of theGm stage per column36μWPower consumption of the EC-ADC per column4.1μWTable 2
Comparison with previous works.
Parameter[21][22][23]a[16]a[24]This workaProcess (nm)1309018018090180Array size580×4501920×1440—1024×768960×720640×480Frame rate (Hz)6050—303550ROIC power (mW)b5064—1042833ADC typePWM-ΔΣSARTS-SSECTS-SSECResolution (bit)121012141214Conversion time (μs)3713.81043.439.740ADC power per column (μW)71.227.372156.3540.1INL (LSB)5.91/-2.280.81/-0.501.06/-0.84+3.69/-9.835.73/-7.30.92/-0.70ENOB (bits)9.4410.511.2510.7c9.13c12.87FoM1 (fJ/step)d779.5452.1—269.0282.6131.1FoM2 (nJ)e3.190.463—4.391.162.15FoM3 (fJ/step)f3792.8261.9295.6391.3449.9214.3aSimulation results. bROICpower=columnADCpower×thenumberofcolumns+ADCperipheralcircuitpower. cENOB is calculated as log22ADCresolution/ε, where ε is the maximum between |INL| and input-referred RMS noise. dFoM1=ROICpower/arraysize×framerate×2ADCresolution. eFoM2=ROICpower/arraysize×framerate. fFoM3=columnADCpower×conversiontime/2ENOB.


## 5. Conclusion

In order to meet the energy efficiency and image quality of the infrared imagers applied in IoT system, this paper presents a low-power ROIC with column-level EC-ADCs. The proposed circuit can be applied to digital readout application for large-scale sensor arrays, such as uncooled IRFPA and CMOS image sensors. Based on circuit noise analysis, a current-reuse topology has been used as theGm stage to boost the transconductance and improve power-noise efficiency. The EC-ADC adopts a pseudodifferential architecture to reduce distortion. According to the nonlinearity analysis, a passive ping-pong integrator architecture is proposed to improve the linearity through reducing the reset time. Moreover, the comparator is shared between the coarse and fine conversion to reduce circuit complexity and power consumption. The comparator employs a positive feedback circuit to improve speed while reducing static power dissipation. To perform a fair performance comparison with previous works, different kinds of FoM were calculated to evaluate the ROIC power efficiency. According to the postsimulation results, the proposed ROIC achieves an ENOB of 12.87 bits, an input-referred noise of 1.74 μVrms (0.89 LSB), a FoM1 of 131.1 fJ/step, a FoM2 of 2.15 nJ, and a FoM3 of 214.3 fJ/step, which meets the low noise, high accuracy, and power-efficiency requirements in IoT sensor readout applications.



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*Source: 5304613-2022-10-03.xml*