# Modeling, Real-Time Estimation, and Identification of UWB Indoor Wireless Channels

Authors: Mohammed M. Olama; Seddik M. Djouadi; Yanyan Li; Aly Fathy
Journal: International Journal of Antennas and Propagation
 (2013)

---

## Abstract

Stochastic differential equations (SDEs) are used to model ultrawideband (UWB) indoor wireless channels. We show that the impulse responses for time-varying indoor wireless channels can be approximated in a mean-square sense as close as desired by impulse responses that can be realized by SDEs. The state variables represent the inphase and quadrature components of the UWB channel. The expected maximization and extended Kalman filter are employed to recursively identify and estimate the channel parameters and states, respectively, from online received signal strength measured data. Both resolvable and nonresolvable multipath received signals are considered and represented as small-scaled Nakagami fading. The proposed models together with the estimation algorithm are tested using UWB indoor measurement data demonstrating the method’s viability and the results are presented.


---

## Body

## 1. Introduction

Ultrawideband (UWB) communication systems have recently attracted significant interest from both the research community and industry since the Federal Communications Commission (FCC) allowed limited unlicensed operation of UWB devices in the USA [1]. They are commonly defined as systems that have either more than 20% relative bandwidth or more than 500 MHz absolute bandwidth. UWB technology has many benefits, including high data rate, low interference, less sensitivity to multipath fading, low transmit power, and availability of low cost transceivers [2]. Industrial standards such as IEEE 802.15.3a and 802.15.4a have been established in recognizing these developments.The ultimate performance limits of a communication system are determined by the channel it operates in [3]. Realistic channel models are thus of utmost importance for system design and testing. UWB propagation channels show fundamental differences from conventional (narrowband) ones in many respects [4, 5], and therefore the established (narrowband) channel models cannot be used. A number of UWB channel models have been proposed in the literature. A model for frequency range below 1 GHz is suggested in [6]. A statistical model that is valid for a frequency range from 3 to 10 GHz is proposed in [7] and is accepted by the IEEE 802.15.4a task group as a standard model for evaluation of UWB system proposals. Significant experimental work in office, residential, and industrial environments has been reported in this field such as in [8, 9]. Most of the proposed channel models are based on characterizing the discrete multipath components. Although these models are able to capture the statistics of the channel, they cannot be specified by a finite number of parameters since their impulse responses are general functions of time and space and therefore are not easy to estimate directly from measurements.A necessary and sufficient condition for representing any time-varying (TV) impulse response (IR) in stochastic state-space form is that it is factorizable into the product of two separate functions of time and space [10]. However, in general this is not the case for the IR of wireless channels. We show that the IR of indoor wireless channels can be approximated in the mean-square sense as close as desired by factorizable impulse responses that can be realized by stochastic differential equations (SDEs) in state-space form. In particular, the SDEs are used to model UWB indoor channels and are combined with system identification algorithms to extract various parameters of the channel from received signal measurement data. The expected maximization (EM) and the extended Kalman filter (EKF) are employed in estimating channel parameters as well as the inphase and quadrature components, respectively. The EM and EKF are chosen since they are recursive and therefore can be implemented online. These algorithms have been recently utilized in [11–13] to estimate the channel parameters and states in narrowband environments, and therefore the formulations of these algorithms are not presented in this paper. Experiments are conducted in our UWB laboratory to collect received signal strength measured data, which are used to determine the applicability of the proposed models. These models can be used in the development of a practical channel simulator that replicates wireless channel characteristics and produces outputs that vary in a similar manner to the variations encountered in a real-world UWB channel environment.Recently, there have been several papers on the application of SDEs to modeling propagation phenomena in radar scattering and wireless communications. SDEs have been successfully used to analyzeK-distributed noise in electromagnetic scattering in [14]. Autoregressive stochastic models for the computer simulation of correlated Rayleigh fading processes are investigated in [15]. A first-order stochastic autoregressive model for a flat stationary wireless channel is introduced in [16]. Stochastic channel models based on SDEs for cellular and ad hoc networks have been presented in [12, 17, 18]. Some preliminary results using SDEs to model UWB channels were presented initially in [19]. The advantage of using SDE methods is based on the computational simplicity of the algorithm simply because estimation is done recursively. This means that there is no need to store and process all measurements; rather, at each time step, the estimator is updated using the previous estimator values and the new innovations.The paper is organized as follows. In Section2, the general TV narrowband and UWB indoor wireless channel impulse responses are introduced. In Section 3, we show that the impulse responses for TV indoor wireless channels can be approximated in a mean-square sense as close as desired by impulse responses that can be realized by SDEs. The stochastic UWB channel models are developed in Section 4. In Section 5, experimental setup and numerical results are presented. Finally, Section 6 provides concluding remarks.


## 2. The General Time-Varying Impulse Response for Indoor Wireless Channels

The general TV impulse response (in complex baseband) of an indoor wireless fading channel is typically represented by Saleh-Valenzuela (SV) model given as [20]
(1)C(t;τ)=∑l=1L∑k=1Klakl(t,τ)exp(jϕkl(t,τ))×δ(t-Tl(t)-τkl(t)),
whereC(t;τ)is the impulse response of the channel at timet, due to an impulse applied at timet-τ,akl(t,τ)andϕkl(t,τ)are, respectively, the random TV tap weight and phase of thekth component in thelth cluster,Tl(t)is the delay of thelth cluster,τkl(t)is the delay of thekth multipath component (MPC) relative to thelth cluster arrivalTl(t),δ(·)is the Dirac delta function,Klis the total number of MPCs within thelth cluster, andLis the total number of clusters that can either be assumed fixed [21] or considered to be a random variable [7]. Lets(t)be the transmitted signal; the received signal is then given by
(2)y(t)=∑l=1L∑k=1Klakl(t,τ)exp(jϕkl(t,τ))×s(t-Tl(t)-τkl(t))+v(t),
wherev(t)is the measurement noise process.For narrowband systems, complex Gaussian fading is conventionally used to describe the small-scale fading. More precisely, the equivalent complex baseband representation consists of Rayleigh-distributed amplitude and uniformly distributed phase. This can be related theoretically to the fact that a large number of multipath components fall into each resolvable delay bin, so that the central limit theorem is valid [3]. Therefore,akl(t,τ)andϕkl(t,τ)are statistically independent Rayleigh and uniform (over[0,2π]) random processes, respectively [20].In UWB systems, the central limit theorem is not valid, and a number of alternative amplitude distributions have been proposed in the literature. The most common empirically determined amplitude distribution in many UWB environments is Nakagami distribution, which is observed in [6, 8] and considered in the IEEE 802.15.4a standard [7]. Therefore, in UWB systems,akl(t,τ)andϕkl(t,τ)are statistically independent Nakagami and uniform random processes, respectively. In the next section, the corresponding impulse response withakl(t,τ)andϕkl(t,τ)for the time-varying indoor wireless channels in (1) is approximated in a mean-square sense as close as desired by SDEs.


## 3. Approximating the Time-Varying Impulse Response for Indoor Wireless Channels by SDEs

Now, we want to represent the TV IR in (1) with a stochastic state-space form in order to allow well-developed tools of estimation and identification to be applied to this class of problems. The following theorem states a necessary and sufficient condition for the realization of the TV IR.Theorem 1 (see [10]).
The impulse responseC(t;τ)of a TV system has a stochastic state-space realization if and only if it is factorizable; that is, there exist functionsg(·)andf(·)such that for alltandτ, one has
(3)C(t;τ)=g(t)f(τ).It is readily seen from the expression of the IRC(t;τ)of the indoor wireless channels in (1) that in general it is not factorizable in the form (3) sinceakl(t,τ)andϕkl(t,τ)are arbitrary functions oftandτ. However, one will show that in generalC(t;τ)can be approximated as close as desired by a factorizable IR function.Theorem 2.
In general, the IRC(t;τ)of the indoor wireless channel in (1) can be approximated as close as desired by a factorizable function.Proof.
The IRC(t;τ)of the indoor wireless channel has finite energy; that is,
(4)C(t;τ)∈L2([0,∞)×[0,∞)),
whereL2([0,∞)×[0,∞))is the Hilbert space of square integrable complex valued functions defined on[0,∞)×[0,∞)with the norm
(5)∥f∥22≔∬[0,∞)[0,∞)|f(t;τ)|2dτdt<∞,f(t;τ)∈L2([0,∞)×[0,∞)).
Likewise defineL2([0,∞))as the standard Hilbert space of square integrable complex valued functions defined on[0,∞)under the norm
(6)∥x∥22≔∫0∞|x(t)|2dt,x∈L2([0,∞)).
The spaceL2([0,∞))contains all finite energy signals defined on[0,∞). The IRC(t;τ)of the channel has a finite energy and belongs toL2([0,∞)×[0,∞)); that is,
(7)∥C∥22≔∬[0,∞)[0,∞)|C(t;τ)|2dτdt<∞.
Since the transmitted and received signals are finite energy signals, the IR can be viewed as an integral operator mapping transmitted signals inL2([0,∞))intoL2([0,∞)); that is, ifs∈L2([0,∞)), then
(8)y(t)=∫0∞C(t;τ)s(t-τ)dτ∈L2([0,∞)).
The tensor space
(9)L2([0,∞))⊗L2([0,∞))≔{F(t,τ)∈L2([0,∞)×[0,∞)):F(t,τ)=∑k=1nαk(t)ϕk(τ),αk(t)∈L2([0,∞)),ϕk(τ)∈L2([0,∞)),∀ninteger}
is dense inL2([0,∞)×[0,∞)); that is, for anyε>0, there exist{αk}1n,{ϕk}1n⊂L2([0,∞))such that [22]
(10)μn≔∥C(t;τ)-∑k=1nαk(t)ϕk(τ)∥2≤ε
andμn→n→∞0. This implies that in the∥·∥2-norm,C(t;τ)can be approximated to any desired accuracy by an IR of the form∑k=1nαk(t)ϕk(τ), which is factorizable by puttingg(t)≔[α1(t)α2(t)⋯αn(t)]andf(τ)≔[ϕ1(τ)ϕ2(τ)⋯ϕn(τ)]T, whereTdenotes vector or matrix transpose.
The optimal approximation ofC(t;τ)by functions inL2([0,∞))⊗L2([0,∞))corresponding to (10) can be written as
(11)μn*≔infαk(t)∈L2([0,∞)),ϕk(τ)∈L2([0,∞))∥C(t;τ)-∑k=1nαk(t)ϕk(τ)∥2.
For arbitraryn, expression (11) is nothing but the shortest distance between the impulse functionC(t;τ)and the spaceL2([0,∞))⊗L2([0,∞)); that is,μn*=dist(C(t;τ),L2([0,∞))⊗L2([0,∞))). The problem is to find the minimizing functions{αk}1n,{ϕk}1n. This problem has been solved in [22] for arbitrary positiven. However, for fixed positiven, the problem becomes the shortest distance, denoted byμno≔dist(C(t;τ),S), fromC(t;τ)to the set
(12)S≔{F(t,τ)=∑k=1nαk(t)ϕk(τ),αk(t)∈L2([0,∞)),ϕk(τ)∈L2([0,∞)),n>0fixedF(t,τ)=∑k=1nαk(t)ϕk(τ)}.
Note thatSis not a subspace and is not a convex set since it is not closed under addition. Therefore, the argument presented in [22] does not hold anymore since the orthogonal projection onto the setSis not linear. Following [22], the impulse response is viewed as an integral operatorTmapping transmitted signals fromL2([0,∞))intoL2([0,∞)); that is, ifsl∈L2([0,∞)), thenyl(t)∈L2([0,∞)), where
(13)yl(t)=(Tsl)(t)≔∫0∞C(t;τ)sl(t-τ)dτ.
Since the impulse response is finite energy, the operatorTis a Hilbert-Schmidt or a trace class 2 operator [22, 23]. Let us denote the class of Hilbert-Schmidt operators acting fromL2([0,∞))intoL2([0,∞))byC2and the Hilbert-Schmidt norm∥·∥HSis defined by
(14)∥T∥HS=∬[0,∞)[0,∞)|C(t;τ)|2dτdt,T∈C2.
The operatorTadmits a spectral factorization of the form [22, 23]
(15)T=∑i=1∞λiνi⊗ψi,
where⊗is the tensor product,λi>0withλi≥λi+1,i=1,2,…,and both{νi}1∞and{ψi}i=1∞are orthonormal sequences inL2([0,∞))and are given by
(16)νi(t)=∫0∞C(t;τ)ψi(τ)dτ,ψi(τ)=∫0∞C(t;τ)νi(t)dt.
The sum (15) has either a finite or countably infinite number of terms. The above representation is unique. The Hilbert-Schmidt norm ofTis also given by
(17)∥T∥HS=∑iλi2<∞.
The spectral factorization (15) yields the following representation for the impulse responseC(t;τ)[23]:
(18)C(t;τ)=∑i=1∞λiνi(t)ψi(τ).
It follows that the minimum in (11) is given by taking. 							(19)αk(t)=λkνk(t),ϕk(τ)=ψk(τ),k=1,2,…,n.
To further illustrate this result, note that{νi}i=1∞and{ψi}i=1∞are orthonormal systems, and simple computations yield
(20)∥C(t;τ)-∑k=1nλkνk(t)ψk(τ)∥22=∥C(t;τ)∥22-∑k=1nλk2.
Schmidt in [24] showed that for any other functionsfi(t),gi(τ)∈L2([0,∞)),i=1,2,…,n, the following inequality holds:
(21)∥C(t;τ)-∑k=1nfk(t)gk(τ)∥22≥∥C(t;τ)∥22-∑k=1nλk2,
and therefore the minimum in (11) is given by (19). The optimal approximation follows as
(22)μno=∥C(t;τ)-∑k=1nλkνk(t)ψk(τ)∥2=∑k=n+1∞λk2
andμno→n→∞0. That is, by increasingn, the RHS of (22) can be made arbitrarily small. In other words, for large enoughn, the following approximation is optimal in a mean-square sense:
(23)C(t;τ)≈∑i=1nλiνi(t)ψi(τ)
and is factorizable.The corresponding SDE is then given by [10]
(24)dX(t)=f(t)dW(t),y(t)=g(t)X(t),
whereX(t)is the state of the channel andW(t)is the standard Brownian motion. Since state-space realizations of impulse responses are not unique [10], a realization of the following form in terms of the inphase and quadrature components for thekth path within thelth cluster can be used as [12, 17]
(25)dXklI(t)=AklI(t)XklI(t)dt+BklI(t)dWklI(t),Ikl(t)=HklI(t)XklI(t),dXklQ(t)=AklQ(t)XklQ(t)dt+BklQ(t)dWklQ(t),Qkl(t)=HklQ(t)XklQ(t),
whereIkl(t)andQkl(t)are, respectively, the inphase and quadrature component processes,XklI(t)andXklQ(t)are, respectively, the state vectors of the inphase and quadrature components,{WklI(t)}t≥0and{WklQ(t)}t≥0are two independent standard Brownian motions which correspond to the inphase and quadrature components, respectively, andAklI(t),AklQ(t),BklI(t),BklQ(t),HklI(t), andHklQ(t)are matrices of appropriate dimensions. Note thatIkl(t)andQkl(t)are two independent Gaussian processes with zero-mean and equal variances; that is,Var(Ikl(t))=Var(Qkl(t))=Var(Nkl(t)), whereNkl(t)is eitherIkl(t)orQkl(t)[13]. They are related to the tap weightakl(t,τ)and phaseϕkl(t,τ)by the expressionsakl(t,τ)=Ikl2(t,τ)+Qkl2(t,τ)andϕkl(t,τ)=arctan(Qkl(t,τ)/Ikl(t,τ)), respectively. In the next section, the stochastic narrowband and UWB indoor channel models are developed.


## 4. Stochastic State-Space Models for UWB Indoor Wireless Channels

In narrowband systems, the equivalent indoor complex baseband representation consists of Rayleigh-distributed amplitude and uniformly distributed phase [20]. Thus, the stochastic narrowband indoor state-space channel model can be represented as
(26)dXkl(t)=Akl(t)Xkl(t)dt+Bkl(t)dWkl(t),y(t)=∑l=1L∑k=1Kl(CklI(t)Xkl(t))2+(CklQ(t)Xkl(t))2×ejarctan(CklQ(t)Xkl(t)/CklI(t)Xkl(t))×s(t-Tl(t)-τkl(t))+D(t)v(t),
where
(27)Xkl(t)=[XklI(t)TXklQ(t)T]T,Wkl(t)=[WklI(t)TWklQ(t)T]T,Akl(t)=[AklI(t)00AklQ(t)],Bkl(t)=[BklI(t)00BklQ(t)],CklI(t)=[HklI(t)0],CklQ(t)=[0HklQ(t)],Ikl(t,τ)=CklI(t)Xkl(t),Qkl(t,τ)=CklQ(t)Xkl(t),
wherev(t)is the measurement noise which is assumed to be Gaussian with zero-mean and unit variance and the tap weight processakl(t,τ)=(CklI(t)Xkl(t))2+(CklQ(t)Xkl(t))2and the phase processϕkl(t,τ)=arctan(CklQ(t)Xkl(t)/CklI(t)Xkl(t))are independent Rayleigh- and uniform-distributed random processes, respectively. In this case, the tap weight processakl(t,τ)has the following statistics [25]:
(28)E{akl(t,τ)}=π2Var(Nkl(t)),E{(akl(t,τ))p}=(2Var(Nkl(t)))p/2Γ(1+12p),Var{akl(t,τ)}=(2-π2)Var(Nkl(t)),
whereΓ(p)is the gamma function.In UWB systems, as mentioned earlier, the most common amplitude distribution of the received signal is Nakagami distribution [6–8]. Its probability density function is given by [26]
(29)f(x)=2Γ(m)(mΩ)mx2m-1exp(-mΩx2),
wherem≥0.5is the shape parameter,Γ(m)is the Gamma function, andΩcontrols the spread of distribution. Them-parameter is often modeled as a random variable [6]. For integer value ofm, the distribution describesmorthogonal independent Rayleigh-distributed random variables. That is, forMRayleigh-distributed random variablesZi, the probability density function of random variableY, defined asY=∑i=1MZi2, is given by a Nakagami distribution with parameterm=M[27].Since multiple orthogonal independent Rayleigh-distributed random variables can generate Nakagami distribution, the stochastic UWB indoor state-space channel model can be represented by(30)dXikl(t)=Aikl(t)Xikl(t)dt+Bikl(t)dWikl(t),y(t)=∑l=1L∑k=1Kl∑i=1Mkl(CiklI(t)Xikl(t))2+(CiklQ(t)Xikl(t))2×ejarctan(CiklQ(t)Xikl(t)/CiklI(t)Xikl(t))×s(t-Tl(t)-τkl(t))+D(t)v(t),
where
(31)Xikl(t)=[XiklI(t)TXiklQ(t)T]T,Wikl(t)=[WiklI(t)TWiklQ(t)T]T,Aikl(t)=[AiklI(t)00AiklQ(t)],Bikl(t)=[BiklI(t)00BiklQ(t)],CiklI(t)=[HiklI(t)0],CiklQ(t)=[0HiklQ(t)],Iikl(t,τ)=CiklI(t)Xikl(t),Qikl(t,τ)=CiklQ(t)Xikl(t),
whereLis the total number of clusters,Klis the total number of resolvable delay bins (paths), andMklis the number of nonresolvable delay bins within the resolvable delay binkin clusterl. In this case, the tap weight processakl(t,τ)has the following statistics [25]:
(32)E{akl(t,τ)}=2Var(Nkl(t))Γ((1/2)(2Mkl+1))Γ(Mkl),E{(akl(t,τ))p}=(2Var(Nkl(t)))p/2Γ((1/2)(2Mkl+p))Γ(Mkl),p≥0,Ωkl=2Var(Nkl(t))Γ(Mkl+1)Γ(Mkl).It can be noticed in (30) that the received signal measurement,y(t), is a nonlinear function of the state variables of the model. The UWB channel parameters are estimated using the EM algorithm, and the inphase and quadrature components are estimated directly from received signal measurements using the EKF. A filter-based EM algorithm together with the EKF is employed to estimate the channel model parameters and states in (30). These filters use only the first- and second-order statistics and are also recursive and therefore can be implemented online. These algorithms have been recently utilized in [11–13] to estimate the channel parameters and states in narrowband environments, and therefore the formulations of these algorithms are not presented in this paper. Experimental results demonstrating the applicability of these algorithms in UWB indoor environments are discussed in the next section.


## 5. Experimental Setup and Numerical Results

A simple Gaussian pulse with clean pulse shape and narrow pulse width is chosen as UWB source signal. The experimental setup is similar to the one in [28]. It comprises a 300-picosecond Gaussian pulse (see Figure 1) that modulates a carrier signal centered at 8 GHz and is transmitted through an omnidirectional UWB antenna. Multiple directional Vivaldi subarray receiving antennas are located at distinct locations in an indoor environment to receive the modulated pulse signal. Each received modulated Gaussian pulse is amplified through a low noise amplifier (LNA) and then stored in a multichannel Tektronix TDS8200 sampling oscilloscope.Figure 1
Transmitted signal of a 300-picosecond Gaussian pulse shape.The transmitter is located next to a metal wall. The receiving antenna is placed 10 cm away from the transmitter and then moved along the same direction away from the transmitter for 20 cm, 50 cm, 1 m, and 2 m. This scenario is demonstrated in Figure2. In this environment, the transmitted signal suffers from reflection from the metal wall, ceiling, or floor or is scattered by the corner of the metal table, and then it reaches the receiver. Figure 3 shows the measured received signal which consists of three clusters. The measurement data are amplified by 30 dB and it can be seen in Figure 4 that the dominant path in each cluster is best fit to Nakagami distribution with different parameters. For example, the dominant path in the first cluster is best fit to Nakagami distribution with parametersm=3andΩ= 0.28. This indicates that there are three nonresolvable delay bins within this path (i.e.,M11= 3). Figure 5 shows the measured and estimated received signals using the EM algorithm together with the EKF for the three clusters using a 4th-order model. It can be noticed that the UWB received signal has been estimated with very high accuracy and it takes little iteration for the estimation algorithm to converge. At a certain time instant, the system parameters for the dominant path in the 1st cluster, which consists of three nonresolvable delay bins, are estimated as follows:
(33)A^111=[01000.19280.0556000001000.1484-0.0966],A^211=[01000.29170.0556000001000.3563-0.0682],A^311=[01000.2014-0.0586000001000.1881-0.2669],B^1112=[0.16990.24510.01220.00090.00040.09610.0287-0.00040.01220.00870.16280.02130.00770.02540.2959-0.0066],B^2112=[0.00060.01820.14290.00770.0043-0.02560.0076-0.25970.07950.00560.00430.0218-0.01770.01920.1541-0.1632],B^3112=[0.25610.00130.33100.00690.01290.0213-0.01070.09210.00440.00150.08240.00760.15410.14290.1632-0.3334],C^111I=C^211I=C^311I=[1000],C^111Q=C^211Q=C^311Q=[0010],D^2=[0.8765].Figure 2
The indoor environment considered in our experiment.Figure 3
The measured UWB received signal.Histogram of measurement data for the dominant paths within the three clusters that are best fit to Nakagami distributions.
(a)
(b)
(c)Measured and estimated received signals using the EM algorithm combined with the EKF for the 1st, 2nd, and 3rd clusters.
(a)
(b)
(c)Notice that the EM algorithm estimatesB^2andD^2instead ofB^andD^. Also note that the parametersL,Kl,Mkl,Tl, andτklare assumed known through the estimation process. In fact, the parametersL,Kl, andMklcan be estimated by various estimation algorithms such as minimum description length [29], multiple hypothesis testing [30], exponential fitting test [31], and linear piecewise variation [32]. And a sounding device is usually dedicated to estimate the time delay of each discrete path (i.e.,Tlandτkl) such as the rake receiver [33].


## 6. Conclusion

This paper describes a general scheme for extracting mathematical UWB indoor channel models from noisy received signal measurements. The UWB channel models are represented in stochastic state-space form, in which its system output produces Nakagami-distributed received signal strength and it is shown to approximate the general TV IR of the channel as close as desired. Experimental results indicate that the measured data can be regenerated with high accuracy.



---

*Source: 467670-2013-12-02.xml*