# UWB Directive Triangular Patch Antenna

Authors: A. C. Lepage; X. Begaud; G. Le Ray; A. Sharaiha
Journal: International Journal of Antennas and Propagation
 (2008)

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## Abstract

Compact directive UWB antennas are presented in this paper. We propose an optimization of the F-probe fed triangular patch antenna. The new design achieves an impedance bandwidth of69% (3–6.15 GHz) and presents good radiation characteristics over the whole impedance bandwidth. The average gain is 6.1 dB. A time-domain study has been performed to characterize the antenna behavior in case a UWB pulse is used. Finally, we propose an alternative solution to facilitate the manufacturing process using metallized foam technology. It also improves the robustness of the antenna as well as reducing its cost.


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## Body

## 1. Introduction

One of the key  issues  of the emerging ultra-wide band (UWB) technology is the design of low cost
compact antennas.Many researches are focused on omnidirectional antennas because their
wide beamwidth enables to communicate with radio elements whatever their
position is.But directional antennas also present some interest. In fact, on the
transmitter side, even if regulation authorities have limited the effective
isotropic-radiated power (EIRP), the use of directional antennas enables to
reduce radiation in undesired directions and thus improves power consumption.
On the receiver side, the antenna gain adds a few and precious dB in the link
budget.Concerning the applications, directional antennas could be placed on a
wall or implemented in a sectorized radio topology (radio-access point) or in a
terminal using multiple directive antennas (antenna diversity). It is also
possible for some specific applications that the user points the antenna in the
desired direction, for example, for a fast download between his terminal and a
PC.With their low thickness, conformability, compactness and low cost,
microstrip antennas are very attractive for such applications, but they suffer
from their inherently narrow bandwidth. In order to solve this problem, a
well-known technique consists in using low permittivity and thick substrate.The extreme case is using the air as the substrate. This solution will
maximize the size of the structure. However, it remains attractive for some
applications as the antenna is still compact.It has been demonstrated in [1] that increasing the thickness of the
substrate induces an inductive effect on input impedance which can be offset by
adding a capacitance to obtain a wide-band behavior.Thus, the L-shaped probe proposed in [2] to feed the patch by
electromagnetic coupling enables to widen the bandwidth. A triangular patch fed
by an L-shaped probe achieves 42% impedance bandwidth and a maximum gain of
6 dBi [3]. An F-shaped probe rectangular patch antenna with an impedance
bandwidth of 64% has also been designed in [4] but the radiation pattern was
not stable over the bandwidth due to the excitation of successive radiating
modes of the patch.In [5], we have proposed a triangular patch fed by an F-shaped probe.
This new structure has stable radiation pattern over the impedance bandwidth of
47%, from 3.1 to 5 GHz.In this paper, we present an improvement of this antenna with a wider
bandwidth.We also propose an alternative solution to manufacture the probe and the patch using metallized
foam technology in order to improve its reproducibility and reduce its cost.


## 2. Description of the F-Probe Antenna

The geometry of the antenna is presented in Figure1. The F-probe is made of a bended 0.2 mm thick metallic rectangular strip on which a second strip has been welded. The whole is welded
to an SMA connector whose central conductor goes through the square ground plane.Geometry of the antenna and prototype.
(a)(b)(c)(d)The radiating element is an isosceles triangle cut out of a steel sheet which is
electromagnetically fed by the F-probe. The triangular shape has been chosen
because it has been demonstrated in [6] that three of the five first modes (TM10, TM20, and TM21) present similar radiation pattern,
polarization, and input impedance. The triangle is centered above the ground
plane and supported by a foam layer; it has been verified to have a
permittivity close to the air.Dimensions of the structure are as follows:Lp = 36 mm, D =
0.6 mm, H3 = 0.4 mm, α = 84°, H = 14.9 mm, H1 = 10.05 mm, H2 = 5.6 mm, L1 = 9.9 mm, L2 = 10.8 mm, w = 3.6 mm, horizontal strips width = 1.2 mm, ground plane size = 67 × 67 mm, probe thickness = 0.4 mm.The antenna has been simulated with CST Microwave Studio software.This optimized structure is the result of a parametric study which has
demonstrated that it is possible to generate multiple resonances close to each
other when the lengths of the horizontal strips of the probe are slightly
different.However, a capacitance effect caused mismatching. That is why we added an inductive
effect on the probe by increasing the width of the vertical strip. Finally, the
optimization has also been done onH, H1, and
H2 to obtain a good impedance matching.This new probe is different from the one presented in [5] whose strips had all the same width and very different lengths.At the beginning of the study, the angleα value was 90°. But due to
constraints concerning the length of the base of the triangle, the value of α has been reduced
to 84°.


## 3. Results and Analysis

### 3.1. Frequency-Domain Results

The simulated and measured return loss is presented in Figure2. The measured
bandwidth, defined for a reflection coefficient |S11| lower than −10 dB,
reaches 69% (3.01–6.15 GHz). Good agreement between simulation and measurement is
observed. And it has been demonstrated that the ground plane size has no
influence on impedance matching. Figure 3 presents measured
radiation pattern at different frequencies over the bandwidth in
the E-plane (XZ plane) and Figure 4 in the H-plane (YZ plane).Figure 2
Measured and simulated reflection coefficient magnitude|S11| versus frequency.Radiation pattern in the E-plane.
(a)(b)Radiation pattern in the H-plane.
(a)(b)Main lobe direction remains very stable in the H-plane and quite stable
in the E-plane with a maximum variation of 15 degrees.Cross-polarization level is very low in the E-plane and in the main lobe
direction of the H-plane. In the H-plane, it is maximum (−6.7 dB) outside the
main lobe forθ = ±82° at 6.15 GHz.Measured and simulated gains in the main lobe direction are presented in
Figure5.Figure 5
Comparison between simulated and measured gain in the main lobe direction over the bandwidth.Measured gain is quite stable up to 5.5 GHz and decreases at higher
frequencies. Average measured gain is 6.1 dB.Good agreement between simulation and measurement is observed except for
highest frequencies, which requires new measurements.The ground plane size influence on gain has also been studied.Figure6 presents the gain in the main lobe direction (θ = 0°, φ = 0°) as a function of the ground plane size
at different frequencies. It can be deduced that a ground plane size of λ0 by λ0 gives the smallest variation of the gain and the best average value over the bandwidth (λ0 is the wavelength
at f = 4.58 GHz). That is why we chose this size for the prototype.Figure 6
Gain in the main lobe direction as a function of the ground plane size at different frequencies. (λ0 is the wavelength corresponding to the central frequency of the impedance bandwidth of the antenna).


### 3.2. UWB Specific Time- and Frequency-Domain Analyses

Evaluation  of input impedance,
radiation pattern, and gain as a function of frequency is not sufficient in
case of using time-domain modulation schemes, for example, pulse position
modulation. The distortion of the pulse resulting from the dispersive nature of
the antenna has also to be analyzed.For this purpose, we measured the transfer function as a function of the angular
coordinates using a reference antenna as described in [7].Once it is determined, it is possible to calculate the radiated field.We chose a Gaussian excitation impulse whose spectrum is presented in Figure7. Figure 8 presents the excitation signal and the measured and simulated radiating field
in the main lobe direction.Figure 7
Spectrum of the excitation signal.Figure 8
Excitation signal and radiated field in the main lobe direction.From Figure8, we can observe very good agreement between simulation
with CST Microwave Studio and the measurement. Although we remark some distortion,
it seems to be moderate: the radiated pulse is not very different from the
excitation signal. It is a bit larger due to the fact that the antenna has
filtered all frequencies outside its impedance bandwidth.Figure9 shows the evolution of the far-field phase (copolarization component) over
the bandwidth for different values of elevation angle in the main lobe in the
E- and H-planes. It is nearly linear which confirms low pulse distortion.Simulated phase of the far-field (copolarization) in the E-plane (left) and in the H-plane
at different angles in the main lobe.
(a)(b)Figure 10
Group delay in the main lobe directionθ = 0° between 2 and 12 GHz.In order to quantify the phase linearity, we studied the group delayτg,
defined in [8] as:
τg=-(∂φ/2π∂f), where φ is the phase and f the frequency.And the standard deviation of the group delay defined in [8] as
σgd=(1/Δf)∫f1f2(τg−τg¯)2df, where f1 and f2 are the boundaries of the impedance bandwidth Δf and τg¯ is the average group delay.For the F-probe antenna, the standard deviation is under 250 ps over the beamwidth.


## 3.1. Frequency-Domain Results

The simulated and measured return loss is presented in Figure2. The measured
bandwidth, defined for a reflection coefficient |S11| lower than −10 dB,
reaches 69% (3.01–6.15 GHz). Good agreement between simulation and measurement is
observed. And it has been demonstrated that the ground plane size has no
influence on impedance matching. Figure 3 presents measured
radiation pattern at different frequencies over the bandwidth in
the E-plane (XZ plane) and Figure 4 in the H-plane (YZ plane).Figure 2
Measured and simulated reflection coefficient magnitude|S11| versus frequency.Radiation pattern in the E-plane.
(a)(b)Radiation pattern in the H-plane.
(a)(b)Main lobe direction remains very stable in the H-plane and quite stable
in the E-plane with a maximum variation of 15 degrees.Cross-polarization level is very low in the E-plane and in the main lobe
direction of the H-plane. In the H-plane, it is maximum (−6.7 dB) outside the
main lobe forθ = ±82° at 6.15 GHz.Measured and simulated gains in the main lobe direction are presented in
Figure5.Figure 5
Comparison between simulated and measured gain in the main lobe direction over the bandwidth.Measured gain is quite stable up to 5.5 GHz and decreases at higher
frequencies. Average measured gain is 6.1 dB.Good agreement between simulation and measurement is observed except for
highest frequencies, which requires new measurements.The ground plane size influence on gain has also been studied.Figure6 presents the gain in the main lobe direction (θ = 0°, φ = 0°) as a function of the ground plane size
at different frequencies. It can be deduced that a ground plane size of λ0 by λ0 gives the smallest variation of the gain and the best average value over the bandwidth (λ0 is the wavelength
at f = 4.58 GHz). That is why we chose this size for the prototype.Figure 6
Gain in the main lobe direction as a function of the ground plane size at different frequencies. (λ0 is the wavelength corresponding to the central frequency of the impedance bandwidth of the antenna).


## 3.2. UWB Specific Time- and Frequency-Domain Analyses

Evaluation  of input impedance,
radiation pattern, and gain as a function of frequency is not sufficient in
case of using time-domain modulation schemes, for example, pulse position
modulation. The distortion of the pulse resulting from the dispersive nature of
the antenna has also to be analyzed.For this purpose, we measured the transfer function as a function of the angular
coordinates using a reference antenna as described in [7].Once it is determined, it is possible to calculate the radiated field.We chose a Gaussian excitation impulse whose spectrum is presented in Figure7. Figure 8 presents the excitation signal and the measured and simulated radiating field
in the main lobe direction.Figure 7
Spectrum of the excitation signal.Figure 8
Excitation signal and radiated field in the main lobe direction.From Figure8, we can observe very good agreement between simulation
with CST Microwave Studio and the measurement. Although we remark some distortion,
it seems to be moderate: the radiated pulse is not very different from the
excitation signal. It is a bit larger due to the fact that the antenna has
filtered all frequencies outside its impedance bandwidth.Figure9 shows the evolution of the far-field phase (copolarization component) over
the bandwidth for different values of elevation angle in the main lobe in the
E- and H-planes. It is nearly linear which confirms low pulse distortion.Simulated phase of the far-field (copolarization) in the E-plane (left) and in the H-plane
at different angles in the main lobe.
(a)(b)Figure 10
Group delay in the main lobe directionθ = 0° between 2 and 12 GHz.In order to quantify the phase linearity, we studied the group delayτg,
defined in [8] as:
τg=-(∂φ/2π∂f), where φ is the phase and f the frequency.And the standard deviation of the group delay defined in [8] as
σgd=(1/Δf)∫f1f2(τg−τg¯)2df, where f1 and f2 are the boundaries of the impedance bandwidth Δf and τg¯ is the average group delay.For the F-probe antenna, the standard deviation is under 250 ps over the beamwidth.


## 4. F-Probe Antennas Using Metallized Foam Technology

The F-probe triangular patch antenna presents interesting features. But one critical issue still
persists: the difficulty of realization which is directly linked to the cost.
In fact, the manufacturing of the F-probe is a real technological issue,
especially the welding of the lower horizontal strip to the bended strip.
Moreover, the robustness of the antenna has to be improved.We retained an innovative solution developed at IETR (Institut
d’Electronique et des Télécommunications de Rennes) which enables to metallize
low-permittivity foam of arbitrary geometry [9].Thus, we realized the antenna with a quite rigid foam whose electrical
permittivity is 1.23.Due to this new permittivity value, it has been necessary to resize the
antenna. New dimensions are as follows:Lp = 32.5 mm, D = 0.6 mm, α = 84°, H = 13.9 mm, H1 = 9.7 mm, H2 = 5.3 mm, L1 = 9.1 mm, L2 = 9.9 mm, w = 3.25 mm, horizontal strips width = 1.1 mm, ground plane
size = 54 × 54 mm, metallization thickness = 0.02 mm.The radiating patch and the F-probe are metallized on
3 different blocks of foam. (Figures11 and 12): the triangle is metallized on
the block n°1, the upper vertical part of the probe and the upper horizontal
strip are on the block n°2 and the lower vertical part and the lower strip are
metallized on the block n°3. As before, the ground plane is cut in a brass
sheet. The SMA connector is welded to the vertical metallized part directly on
the foam. The three blocks are screwed together and also to the ground plane
with 4 nylon screws. (Figure 13).Figure 11
Structure of the  antenna using metallized foam technology.Figure 12
Metallized foam blocks before assembly.Figure 13
Triangular patch F-probe antenna using metallized foam technology.In order to evaluate the reproducibility of the process, we realized two prototypes.Figure14 presents the measurement of the magnitude of the reflection
coefficient for the two prototypes and the comparison with the simulation result.Figure 14
Measured and simulated reflection coefficient magnitude|S11| versus frequency.First of all, results are similar for the two prototypes which demonstrate the good reproducibility of
the manufacturing process.The measured bandwidth reaches 58% (3.2–5.8 GHz).The agreement between measurement and simulation is satisfying.Due to space constraints, radiation patterns are not shown but they
appear to be similar to those presented in Section3.Since the relative electric permittivity is 1.23, the gain value is
similar to the one of the previous design.The next step of improvement of the manufacturing of this antenna will be the metallization of the
ground plane.


## 5. Conclusion

In this article, we have presented an optimization performed on the
F-probe triangular patch antenna. The new design achieves an impedance
bandwidth of 69% with a stable radiation pattern over the bandwidth and is
quite compact (λ0 × λ0 × 0.22λ0). A time domain
study has also shown that the  antenna distorts the excitation pulse in a moderate way.A solution has also been proposed for an easier realization and a better
robustness of the probe in the purpose of minimizing manufacturing costs and improving reproducibility.At this time, this antenna is integrated in an
industrial UWB test platform.



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*Source: 410786-2008-02-03.xml*