# Codesigned Wideband High-Efficiency Filtering SIW Slot Antenna with High Selectivity and Flat Gain Response

Authors: Peng Kai Li; Chang Jiang You; Hong Fang Yu; Yu Jian Cheng; Yuan Wang Yang; Jian Hua Deng
Journal: International Journal of Antennas and Propagation
 (2018)

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## Abstract

A substrate integrated waveguide (SIW) slot antenna with wideband bandpass filtering performance is proposed in this journal. It consists of a SIW cavity, a transverse slot, three metal posts, and a SMA connector. Three metal posts split the SIW cavity into two TE110-mode resonators. A long nonresonant transverse slot is utilized to realize a half TE110-mode resonator and generate radiation simultaneously so as to reduce the size and release the use of extra radiator. Three in-band resonance poles and a radiation null at both out-of-bands are obtained. A prototype is fabricated and measured. Measured results demonstrate the proposed antenna is with a center frequency of 4.26 GHz, a fractional bandwidth of 9.1%, a high efficiency of 93%, and flat gain response and good skirt selectivity of 270 dB/GHz and 330 dB/GHz for the lower and upper out-of-band, respectively.


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## Body

## 1. Introduction

Modern wireless communications demand the RF frontend system to be compact, lightweight, low cost, high efficiency, and multifunctional. In most of the RF front ends, the bandpass filter and the antenna are the key components whose performance will immediately affect the system performance. In the traditional systems, they are usually designed separately and connected by a 50 Ω or 75 Ω transmission line, which not only increases the volume but also maybe degrades in-band performance due to the mismatch and extra insertion loss caused by the interconnections [1]. Recently, a concept of filtering antenna has been proposed by integrating the bandpass filter and the antenna into a single component with filtering and radiating functions simultaneously. The proper integration of the antenna and filter turns out to be an efficient way to reduce the loss and enhance the efficiency of this functional block for a frontend system [2, 3].Substrate integrated waveguide (SIW) technology has been effectively applied to design high-performance filters and antennas due to its low insertion loss [4–6]. Some filtering antennas based on SIW technology have been reported in [7–12]. In [7, 8], the antenna is planar and designed by cascading resonators and radiator. In [9–12], 3D configurations by placing the resonators under the radiator are applied. However, in reported works, they suffer from more insertion loss as they all need an extra radiator to cascade the filtering circuit and the filtering circuit is realized by full mode resonators which will lead to a large lossy circuit size. What is more, as they have no radiation null (transmission zero) in gain response or the radiation nulls are far away from the pass-band, the out-of-band skirt selectivity in the reported works is low.To overcome these problems, a codesigned wideband filtering substrate integrated waveguide (SIW) slot antenna with high selectivity, high efficiency, and flat gain response is proposed in this journal. A long nonresonant transverse slot is utilized to realize a half mode resonator and generate radiation simultaneously so as to reduce the size and release the use of extra radiator. Three resonance poles are achieved to enhance the bandwidth and gain flatness. Two radiation nulls are introduced to increase the out-of-band skirt selectivity. Antenna work mechanism is explained, and the results and discussion are given.


## 2. Antenna Configuration and Mechanism

### 2.1. Configuration of the Proposed Antenna

Figure1 shows the configuration of the proposed filtering antenna. The antenna is designed on a F4B-2 substrate with a thickness of 6 mm, a relative permittivity of 2.485, and a loss tangent of 0.0018. It is composed of a SIW cavity with size W×L, a transverse slot with size ws×ls etched on the top metal layer, three metal posts in the cavity and a SMA connector as feed. The metal posts are with a diameter of d and space of xt. The antenna configuration is symmetrical about the y-axis. The antenna is designed and optimized at the centre frequency of 4.25 GHz. The detailed dimensions are as follows: L=51, W=49, m=5.3, d=2.4, xt=13.8, l1=23, l2=13, ls=29.4, ws=2.65, a=1.6, and s=3.2 (unit: mm).Figure 1
Configuration of the proposed filtering antenna. (a) Top view. (b) Side view.
(a)
(b)The design evolution of the proposed antenna is illustrated in Figure2. Firstly, a transverse slot is etched on the upper metal layer of SIW with end shorted (Ant. 1). Then, three metal posts are introduced at the right side of the slot (Ant. 2). Finally, a SIW cavity is formed and a SMA connector is used as feed (Ant. 3). It should be mentioned that the slot is nonresonant at the work band and its length is more than half a guide wavelength. The reflection coefficient S11 and peak realized gain of Ant. 1–3 are given in Figure 3. From Figure 3(a), it can be found that the proposed antenna Ant. 3 has three resonance poles fr1, fr2, and fr3. From Figure 3(b), it can be found that Ant. 1 has only one radiation null in gain response while Ant. 2 and Ant. 3 have two radiation nulls at almost the same frequencies. These phenomena will be explained in detail in the following section.Figure 2
Design evolution. (a) SIW transverse slot. (b) SIW transverse slot with metal posts. (c) SIW cavity-backed transverse slot with metal posts.
(a)
(b)
(c)Figure 3
Simulated results of Ant. 1, Ant. 2, and Ant. 3. (a) Reflection coefficientS11. (b) Peak realized gain.
(a)
(b)


### 2.2. Antenna Mechanism of Radiation Nulls

The configuration of a SIW transmission line with end shorted is shown in Figure4(a). When it is excited by TE10 mode, the surface current I is with standing wave distributions, as expressed by
(1)I=Imaxcosβfz.Figure 4
SIW transmission line with end shorted and surface current distributions. (a) Configuration of SIW transmission line with end shorted. (b) Surface current distributions at different frequencies.
(a)
(b)Here,Imax is defined as the maximum current magnitude, βf denotes the phase constant at frequency f, and z means the distance from the shorted end. Figure 4(b) gives the surface current distribution at two different frequency f1 andf2, which are labelled as case 1 and case 2, respectively. If a slot is etched on the top layer, it will generate radiation or not which depends on the current distribution at the position of the slot. It should be stated that there will be no radiation at the current null. Thus, if the slot is located at the position of current null as case 1 in Figure 4(b), there will be no radiation generated from the slot. And if the slot is not located at the position of current null as case 2, there will be radiation generated from the slot. Hence, a radiation null will occur at the frequency where the current at the slot satisfies the following condition that
(2)IAA′=0.That is the reason why one radiation null is generated in Ant. 1, which is also explained in [13].Actually in Ant. 2 and Ant. 3, when three metal posts are introduced, a resonatorR1 is formed. The configuration of SIW transmission line loaded by a resonator and its equivalent circuit model is illustrated in Figure 5(a). The resonator R1 can be simply modelled as shunt inductor and capacitor, and it resonates at 4.25 GHz. The phase shift at the reference plane in the one end shorted structure is twice that of a two-port resonator model as discussed in [14]. Thus, as shown in Figure 5(b), the phase shift φ1 of the end shorted resonator R1 can be expressed as
(3)φ1≈+180°belowresonance,φ1≈−180°aboveresonance.Figure 5
Illustration of analysis model and the results. (a) Analysis model of an end shorted resonator. (b) Simulated phase ofS11.
(a)
(b)For Ant. 2 and Ant. 3, the resonatorR1 can be considered as the load of the transmission line, as illustrated in Figure 6(a). The phase difference φ at the reference plane can be written as
(4)φ=2βfl2+φ1.Figure 6
Illustration of analysis model and the results. (a) Analysis model of Ant. 2. (b) Simulated phase difference.
(a)
(b)Here,l2 denotes the distance from the three metal posts. According to transmission line theory [15], the current strength at the reference plane can be expressed as
(5)Iz=−l2=V0+Z01−Γe−jφ.Here,V0+ means the voltage of incident wave, Z0 denotes the characteristic impedance of the transmission line, and Γ is voltage reflection coefficient. According to (5), the minimum current strength (current null) satisfies the condition that
(6)φ=2nπ,n=0,1,2….According to (3), (4), and (6), when current null occurs, the minimum l2 should meet that
(7)2βfl2≈π.It means whenl2 is about λg/4 (λg is guide wavelength), two current nulls will occur at both sides of the resonance. Figure 6(b) gives the phase difference when l2 changes near λg/4 (13 mm). The phase difference is modularized by 2π and is limited to the [−180° and 180°] range. It can be found when φ=0, there exist two frequency fn1 and fn2 at both sides of 4.25 GHz. From the abovementioned analysis, the frequency where φ=0 is the frequency of current null. It can also be seen from Figure 6(b), frequencies of the two current nulls shift downward when l2 increases from 11 mm to 15 mm. It means the upper current null will come close to the resonance, and the lower current null will be away from the resonance.According to transmission line theory, input impedance is defined asZin=Vz/Iz. When the current is minimum, the input impedance is maximum. So the current null can be deduced from the feature of input impedance. Figure 7 gives the simulated magnitude of input impedance ZAA′ with the variation of l2. It can be found that one impedance peak is observed at both sides of 4.25 GHz. The frequency of the impedance peak is the frequency of current null. Due to the lossy material used in the simulation, when approaching the resonance, the loss is more and voltage reflection coefficient Γ becomes smaller. Meanwhile, the minimum current strength becomes larger according to (5). It means the impedance peak becomes smaller near the resonance 4.25 GHz as can be seen in Figure 7. So the depth of the two radiation nulls in Ant. 2 and Ant. 3 can be estimated from Figure 7. A larger impedance peak means a deeper radiation null, as can be proved from the simulated peak realized gain of Ant. 3 with different l2 in Figure 8.Figure 7
Simulated magnitude ofZAA′ illustrated in Figure 5(a).Figure 8
Simulated peak realized gain of Ant. 3 with differentl2.To achieve two symmetrical radiation nulls about 4.25 GHz, here in the proposed antenna, the slot is located at the position wherel2 is chosen to be 13 mm which is about a quarter phase wavelength at the resonant frequency of R1.


### 2.3. Antenna Mechanism of Resonance Poles

As can be seen in Figure3(a), three resonance poles fr1, fr2, and fr3 are generated in Ant. 3. Actually, when three metal posts are introduced, the large cavity is divided into two TE110-mode resonators. What’s more, when a long slot is cut, a half TE110-mode resonator is created. The three resonant poles are generated by the couplings between the three resonators, and they can be explained by the basic resonant mode superposition. Figure 9 gives the simulated H-filed at different phase of the three resonance poles and their superposition modes. The modes created by the two TE110-mode resonators and their coupling can be considered as an even mode and an odd mode [16]. Thus, as illustrated in Figure 9(a), the first resonance pole fr1 can be considered as the superposition of a half TE110 mode and an even mode. As illustrated in Figure 9(b), the second resonance pole fr2 can be considered as the superposition of a half TE110 mode and a TE110 mode. As illustrated in Figure 9(c), the third resonance pole fr3 can be considered as the superposition of a half TE110 mode and an odd mode. It should be noticed that all the resonance poles are in relation to the half TE110 mode. This is because only the half TE110 mode can generate radiation through the slot, while at even and odd mode, the slot is almost located at the position of current null. It is innovative that in the proposed design, the slot not only acts as a radiator but also is essential to the formation of the half mode resonator. This merged structure is different to the commonly used method which will introduce extra radiator.Figure 9
Illustration of the mechanism of resonance poles in Ant. 3. (a) First resonance polefr1. (b) Second resonance pole fr2. (c) Third resonance pole fr3.
(a)
(b)
(c)


## 2.1. Configuration of the Proposed Antenna

Figure1 shows the configuration of the proposed filtering antenna. The antenna is designed on a F4B-2 substrate with a thickness of 6 mm, a relative permittivity of 2.485, and a loss tangent of 0.0018. It is composed of a SIW cavity with size W×L, a transverse slot with size ws×ls etched on the top metal layer, three metal posts in the cavity and a SMA connector as feed. The metal posts are with a diameter of d and space of xt. The antenna configuration is symmetrical about the y-axis. The antenna is designed and optimized at the centre frequency of 4.25 GHz. The detailed dimensions are as follows: L=51, W=49, m=5.3, d=2.4, xt=13.8, l1=23, l2=13, ls=29.4, ws=2.65, a=1.6, and s=3.2 (unit: mm).Figure 1
Configuration of the proposed filtering antenna. (a) Top view. (b) Side view.
(a)
(b)The design evolution of the proposed antenna is illustrated in Figure2. Firstly, a transverse slot is etched on the upper metal layer of SIW with end shorted (Ant. 1). Then, three metal posts are introduced at the right side of the slot (Ant. 2). Finally, a SIW cavity is formed and a SMA connector is used as feed (Ant. 3). It should be mentioned that the slot is nonresonant at the work band and its length is more than half a guide wavelength. The reflection coefficient S11 and peak realized gain of Ant. 1–3 are given in Figure 3. From Figure 3(a), it can be found that the proposed antenna Ant. 3 has three resonance poles fr1, fr2, and fr3. From Figure 3(b), it can be found that Ant. 1 has only one radiation null in gain response while Ant. 2 and Ant. 3 have two radiation nulls at almost the same frequencies. These phenomena will be explained in detail in the following section.Figure 2
Design evolution. (a) SIW transverse slot. (b) SIW transverse slot with metal posts. (c) SIW cavity-backed transverse slot with metal posts.
(a)
(b)
(c)Figure 3
Simulated results of Ant. 1, Ant. 2, and Ant. 3. (a) Reflection coefficientS11. (b) Peak realized gain.
(a)
(b)


## 2.2. Antenna Mechanism of Radiation Nulls

The configuration of a SIW transmission line with end shorted is shown in Figure4(a). When it is excited by TE10 mode, the surface current I is with standing wave distributions, as expressed by
(1)I=Imaxcosβfz.Figure 4
SIW transmission line with end shorted and surface current distributions. (a) Configuration of SIW transmission line with end shorted. (b) Surface current distributions at different frequencies.
(a)
(b)Here,Imax is defined as the maximum current magnitude, βf denotes the phase constant at frequency f, and z means the distance from the shorted end. Figure 4(b) gives the surface current distribution at two different frequency f1 andf2, which are labelled as case 1 and case 2, respectively. If a slot is etched on the top layer, it will generate radiation or not which depends on the current distribution at the position of the slot. It should be stated that there will be no radiation at the current null. Thus, if the slot is located at the position of current null as case 1 in Figure 4(b), there will be no radiation generated from the slot. And if the slot is not located at the position of current null as case 2, there will be radiation generated from the slot. Hence, a radiation null will occur at the frequency where the current at the slot satisfies the following condition that
(2)IAA′=0.That is the reason why one radiation null is generated in Ant. 1, which is also explained in [13].Actually in Ant. 2 and Ant. 3, when three metal posts are introduced, a resonatorR1 is formed. The configuration of SIW transmission line loaded by a resonator and its equivalent circuit model is illustrated in Figure 5(a). The resonator R1 can be simply modelled as shunt inductor and capacitor, and it resonates at 4.25 GHz. The phase shift at the reference plane in the one end shorted structure is twice that of a two-port resonator model as discussed in [14]. Thus, as shown in Figure 5(b), the phase shift φ1 of the end shorted resonator R1 can be expressed as
(3)φ1≈+180°belowresonance,φ1≈−180°aboveresonance.Figure 5
Illustration of analysis model and the results. (a) Analysis model of an end shorted resonator. (b) Simulated phase ofS11.
(a)
(b)For Ant. 2 and Ant. 3, the resonatorR1 can be considered as the load of the transmission line, as illustrated in Figure 6(a). The phase difference φ at the reference plane can be written as
(4)φ=2βfl2+φ1.Figure 6
Illustration of analysis model and the results. (a) Analysis model of Ant. 2. (b) Simulated phase difference.
(a)
(b)Here,l2 denotes the distance from the three metal posts. According to transmission line theory [15], the current strength at the reference plane can be expressed as
(5)Iz=−l2=V0+Z01−Γe−jφ.Here,V0+ means the voltage of incident wave, Z0 denotes the characteristic impedance of the transmission line, and Γ is voltage reflection coefficient. According to (5), the minimum current strength (current null) satisfies the condition that
(6)φ=2nπ,n=0,1,2….According to (3), (4), and (6), when current null occurs, the minimum l2 should meet that
(7)2βfl2≈π.It means whenl2 is about λg/4 (λg is guide wavelength), two current nulls will occur at both sides of the resonance. Figure 6(b) gives the phase difference when l2 changes near λg/4 (13 mm). The phase difference is modularized by 2π and is limited to the [−180° and 180°] range. It can be found when φ=0, there exist two frequency fn1 and fn2 at both sides of 4.25 GHz. From the abovementioned analysis, the frequency where φ=0 is the frequency of current null. It can also be seen from Figure 6(b), frequencies of the two current nulls shift downward when l2 increases from 11 mm to 15 mm. It means the upper current null will come close to the resonance, and the lower current null will be away from the resonance.According to transmission line theory, input impedance is defined asZin=Vz/Iz. When the current is minimum, the input impedance is maximum. So the current null can be deduced from the feature of input impedance. Figure 7 gives the simulated magnitude of input impedance ZAA′ with the variation of l2. It can be found that one impedance peak is observed at both sides of 4.25 GHz. The frequency of the impedance peak is the frequency of current null. Due to the lossy material used in the simulation, when approaching the resonance, the loss is more and voltage reflection coefficient Γ becomes smaller. Meanwhile, the minimum current strength becomes larger according to (5). It means the impedance peak becomes smaller near the resonance 4.25 GHz as can be seen in Figure 7. So the depth of the two radiation nulls in Ant. 2 and Ant. 3 can be estimated from Figure 7. A larger impedance peak means a deeper radiation null, as can be proved from the simulated peak realized gain of Ant. 3 with different l2 in Figure 8.Figure 7
Simulated magnitude ofZAA′ illustrated in Figure 5(a).Figure 8
Simulated peak realized gain of Ant. 3 with differentl2.To achieve two symmetrical radiation nulls about 4.25 GHz, here in the proposed antenna, the slot is located at the position wherel2 is chosen to be 13 mm which is about a quarter phase wavelength at the resonant frequency of R1.


## 2.3. Antenna Mechanism of Resonance Poles

As can be seen in Figure3(a), three resonance poles fr1, fr2, and fr3 are generated in Ant. 3. Actually, when three metal posts are introduced, the large cavity is divided into two TE110-mode resonators. What’s more, when a long slot is cut, a half TE110-mode resonator is created. The three resonant poles are generated by the couplings between the three resonators, and they can be explained by the basic resonant mode superposition. Figure 9 gives the simulated H-filed at different phase of the three resonance poles and their superposition modes. The modes created by the two TE110-mode resonators and their coupling can be considered as an even mode and an odd mode [16]. Thus, as illustrated in Figure 9(a), the first resonance pole fr1 can be considered as the superposition of a half TE110 mode and an even mode. As illustrated in Figure 9(b), the second resonance pole fr2 can be considered as the superposition of a half TE110 mode and a TE110 mode. As illustrated in Figure 9(c), the third resonance pole fr3 can be considered as the superposition of a half TE110 mode and an odd mode. It should be noticed that all the resonance poles are in relation to the half TE110 mode. This is because only the half TE110 mode can generate radiation through the slot, while at even and odd mode, the slot is almost located at the position of current null. It is innovative that in the proposed design, the slot not only acts as a radiator but also is essential to the formation of the half mode resonator. This merged structure is different to the commonly used method which will introduce extra radiator.Figure 9
Illustration of the mechanism of resonance poles in Ant. 3. (a) First resonance polefr1. (b) Second resonance pole fr2. (c) Third resonance pole fr3.
(a)
(b)
(c)


## 3. Results and Discussion

The proposed antenna is fabricated and measured, and the photograph is shown in Figure10. The simulated and measured reflection coefficient S11, realized gains, and total efficiencies are illustrated in Figure 11. It can be found that measured bandwidth (S11<−10dB) is 9.1% (4.09–4.48 GHz), agreeing well with the simulated one of 10.1% (4.05–4.48 GHz).Figure 10
Photograph of the fabricated antenna.Figure 11
Simulated (dashed lines) and measured (solid lines) reflection coefficientS11, realized gains, and total efficiencies.The measured and simulated average realized gains over the work band are 6.2 dBi and 6.3 dBi, respectively. The gain response is flat over the work band with ripples less than 0.4 dB, which is attributed to the nonresonant slot radiation. The bandpass filtering performance is remarkable, as the efficiency over the work band reaches to a maximum as high as 93% which means insertion loss of about only 0.3 dB is introduced, while the out-of-band efficiency approaches zero. The high efficiency is benefited from the nonresonant slot radiation and small lossy circuit area as a half mode resonator is realized and no extra radiator is utilized. As can be observed in Figure11, two measured radiation nulls at 4.04 GHz and 4.56 GHz in gain response are very close to the pass-band, so high skirt selectivity is achieved. The measured lower and upper out-of-band skirt selectivity is of 270 dB/GHz and 330 dB/GHz, respectively.The simulated and measured radiation patterns at the three resonance poles 4.12 GHz, 4.26 GHz, and 4.46 GHz are shown in Figure12. It can be seen almost symmetrical radiation patterns are observed in E/H plane. The slightly tilts at E-plane patterns are mainly caused by the off-centered slot position. Good front-to-back ratios and cross polarization levels of more than 20 dB are obtained at both planes. The simulated copolarization patterns are in accordance with the measured ones, and the copolarization patterns are stable over the work band. The discrepancy in cross polarization between simulated and measured value is mainly attributed to the slightly tilted antenna placement during measurement.Figure 12
Simulated and measured radiation patterns at three resonance poles. (a) 4.12 GHz. (b) 4.26 GHz. (c) 4.46 GHz.
(a)
(b)
(c)A comprehensive comparison with previous works utilizing SIW slot is summarized in Table1. It can be found in our work the advantages of wideband, high selectivity, and high efficiency (Eff.) compared to other works. The high efficiency can also be deduced from the lossy circuit area which can be evaluated by size × layer. It is obvious that our work has the least value of size × layer.Table 1
Comparisons with previous reported works.
Ref
Number of Pole
Number of Null
BW (%)
Selectivity (dB/GHz)
Eff. (%)
Size (λg×λg)
Layer
[8]
4
0
6
113/74
89
1.3 × 1.7
1
[9]
2
0
3
63/28
58.9
0.9 × 0.9
2
[10]
3
2
1.6
96/55
86
0.7 × 0.9
3
[11]
3
0
5.5
26/57
N.A.
1.4 × 1.6
2
Ours
3
2
9.1
270/330
93
1.1 × 1.2
1
λg is the guide wavelength at the centre frequency of the work band. Selectivity is calculated as αmax−αmin/fz−fc, where αmax is the attenuation of the first out-of-band radiation null or 20 dB attenuation if no radiation null and αmin is the 3 dB attenuation; fz and fc are their corresponding frequency.


## 4. Conclusion

A substrate integrated waveguide slot antenna with bandpass filtering performance in gain response is presented in this journal. A long nonresonant slot is introduced to realize a half mode resonator and generate radiation so as to reduce the size and release the use of extra radiator. Two radiation nulls are generated to enhance the selectivity. The measured results show that wide bandwidth of 9.1%, high selectivity of 270/330 dB/GHz for lower/upper out-of-band, and high efficiency of 93% and flat gain response are obtained in the proposed antenna. The high performance indicates it is a good candidate for the functional module integrated with filter and antenna in RF front system.



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*Source: 3949675-2018-07-02.xml*