# High Gain Superstrate Loaded Membrane Antenna Based on Substrate Integrated Waveguide Technology

Authors: Hamsakutty Vettikalladi
Journal: International Journal of Antennas and Propagation
 (2015)

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## Abstract

The design and the results of a single slot coupled substrate integrated waveguide (SIW) fed membrane antenna loaded with a superstrate layer are presented for 94 GHz communication system. The membrane antenna is designed using ANSYS HFSS and consists of 6 layers. The microstrip patch antenna (MPA) placed on the top pyralux substrate layer is excited by means of a longitudinal rectangular slot placed over the SIW structure in the bottom pyralux substrate. The simulated antenna impedance bandwidth is found to be 5 GHz (91.5–96.5 GHz) with a gain of 7 dBi. In order to improve the gain a superstrate layer is added above the membrane antenna. The maximum gain achieved is 14.4 dBi with an efficiency of 77.6% at 94 GHz. The results are verified using CST Microwave Studio and are found to be in good agreement.


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## Body

## 1. Introduction

During the last decade significant advances in millimeter wave technologies have been made to cope with the increasing interest. The millimeter-wave (mmW) technology has gained a lot of attention in recent years. The W-band (75–110 GHz) window centered at 94 GHz is in focus due to its unique property of high transmission through atmospheric barriers like smoke, thin dielectrics, and clouds [1] as well as for the development of high resolution imaging applications and ultra broadband wireless communications [2]. Furthermore, the shorter wavelength at mmW permits the production of compact systems for various advance communication systems such as remote sensing, radio astronomy, cloud radar, automotive collision warning, and multi gigabits per second point to point communications.The general requirements for mmW antennas concerns wide/ultrawide impedance bandwidth, high radiation efficiency, high gains, and compatibility with other RF communications modules. One of the suitable technologies for mmW antennas is substrate integrated circuits (SICs) [3–5]. In principle the SICs are used to synthesize and convert nonplanar structures and 3D geometries into planar form which makes SICs very attractive for millimeter wave applications [6]. Any nonplanar structure can be converted to its equivalent planar structure utilizing SICs technologies such as substrate integrated waveguide (SIW) [2]. The substrate integrated waveguide (SIW) [3–5] is the most matured and popular structure in family of SICs. Several advantages of a rectangular waveguide such as high power handling, high Q-factor, and electrical shielding are attained in SIW technology. Furthermore, the radiation leakage can be ignored in the SIW structures having metallic via placed in close proximity, resulting in propagation characteristics similar to metallic rectangular waveguides [7]. The SIW based structures can be implemented by various manufacturing processes such as conventional printed circuit board process (PCB) [8, 9], multilayer PCB process [10], photoimageable thick film technology [11], and low-temperature cofired ceramic (LTCC) technique. There are many techniques to improve the gain, either by using large arrays [12, 13], which in turn increases losses from the feed network especially at the high frequencies, or by using superstrates [14, 15]. This technique is simple and efficient to improve the gain.In this paper we present the design and the results of a superstrate loaded multilayer membrane antenna based on SIW technology operating at 94 GHz. The available low cost, low loss DuPont pyralux TK185018R, and FR4 substrates are utilized in the proposed membrane base antenna structure and a high permittivity RT-duroid 6010ML substrate for the superstrate material. The ANSYS HFSS is utilized for modelling and the optimization of proposed antenna, whereas CST Microwave studio is used for validation of the results.


## 2. SIW Fed Base Membrane Antenna

### 2.1. Design and Configuration

The single element SIW based structure of the proposed membrane antenna with 6 layers is shown in Figure1. The two substrates are utilized in design, that is, pyralux substrate with εr=2.4 and loss tangent tan⁡δ=0.002 and FR4 substrates with εr=4.4 and loss tangent tan⁡δ=0.02. The losses are incorporated in simulation. As shown in Figure 1(a) the top layer consists of a microstrip patch antenna (MPA) etched below the pyralux substrate. The second layer consists of the FR-4 substrate with an air cavity to support the top substrate layer. An air gap is drilled on the second layer to enhance the bandwidth of the antenna. We used the FR4 substrate because it is very cheap, low in loss, and compatible with mass production PCB technology electronics. The bottom three layers makes up a SIW slot antenna, where rectangular slot [16] in the top metallic layer of SIW is utilized to excite the MPA through the air gap made within the FR4 layer. The metallic via holes are made with in the bottom layer of pyralux substrate to form the SIW structure. The proposed membrane antenna structure in XZ plane is shown in Figure 1(b) for a better understanding of various layers. All copper layers used in proposed antenna structure have thickness of 18 μm, while the thickness of pyralux and FR-4 is taken to be 50 μm and 100 μm, respectively.Figure 1
Proposed membrane antenna geometry.
(a)
3D layered model
(b)
2D side viewThe distance between two rows of metallic via holes along with the dielectric between them determines the cutoff frequency. The SIW design generally works inTEn0 mode and does not support propagation of TM modes. For the dielectric filled waveguide (DFW) with same cutoff frequency, the broad side dimension of waveguide, that is, ad, is found by [17](1)ad=aεr,where εr is the dielectric constant of substrate and dimensions of “a” are taken from the standard WR-10 waveguide (i.e., 2.54 mm). Once the dimension ad for DFW is known, we can use the following equation to find the separation distance, that is, as, between the via rows of SIW [18]:(2)as=ad+d20.95p,where d is the diameter of metallic via holes connecting the upper and lower metallic layers of the bottom pyralux substrate. The via diameter (d) is taken by d=λg/5 and pitch, that is, centre to centre distance between the via holes, is taken by p<2d [19].In Figure2 the top view of proposed membrane antenna along with longitudinal slot placed on the top ground plane of SIW structure is shown. Usually the distance from the short circuited end of SIW to the centre of slot, that is, doff⁡2, is a quarter or odd multiple of quarter of the guided wavelength. Initially, this distance is chosen to be three quarter of the guided. Furthermore, the slot offset, that is, doff⁡1, is optimized for proper excitation of longitudinal slot as well as the metallic patch antenna on the top layer. The initial dimension for longitudinal slot length is taken by [17](3)Ls=λo2εr+1.Figure 2
Top view of SIW antenna along with rectangular slot.The pyralux substrate with patch on top layer is supported by a layer of FR4 substrate having an optimized air gap of 1.8 mm × 1.8 mm under the MPA. The optimized dimensions for MPA are found to be 1.12 × 0.99 mm. The total size of antenna is found to be 10 mm × 10 mm. A conventional 50 Ω microstrip line is utilized as a feed element in the proposed antenna. The microstrip to SIW transition consisting of a tapered microstrip is also optimized for proper impedance matching. The detailed view of proposed single membrane antenna along with various important design parameters for MPA and air gap in FR-4 substrate are well explained in [20] and in Figure 3.Figure 3
Top view of proposed membrane antenna geometry.The optimum dimension for the proposed membrane antenna structure isWo=0.145 mm, Wt=1.34 mm, Lo=1 mm, Lt=0.9 mm, d=0.5 mm, p=0.7 mm, as=1.5 mm, Ws=0.2 mm, Ls=1.225 mm, L=3.5 mm, doff⁡1=0.55 mm, doff⁡2=1.4125 mm, La=1.85 mm, Wa=1.85 mm, Lp=1.15 mm, Wp=1.05 mm, L=10 mm, and W=10 mm.


### 2.2. Results and Discussion

The proposed antenna structure is simulated and optimized using ANSYS High Frequency Structure Simulator (HFSS©). The results are further verified by simulating the proposed antenna structure in CST Microwave Studio.A comparison between the simulation results ofS11 and gain obtained from HFSS and CST are given in Figure 4. The two resonances at 92.4 GHz and 95.5 GHz are due to presence of slot and patch in antenna geometry, respectively, which are kept in close proximity to achieve wide impedance bandwidth. The FR-4 support with air gap plays an important role in merging the two resonances to achieve a wide impedance bandwidth. The antenna impedance bandwidth is found to be 5.4 GHz (91.5–96.9 GHz) from HFSS while from CST the impedance bandwidth is found to be approximately 4.9 GHz (91.5–96.4 GHz). The gain is found to be 7 dBi at the center frequency of 94 GHz. The peak gain of 7.5 dB is found at 95.5 GHz. Furthermore, the antenna gain remains above 6.7 dB in whole frequency band of operation.Figure 4
Simulated return lossS11 and realized gain.The antenna radiation pattern in both E-plane (horizontal plane) and H-plane (vertical plane) remains similar throughout the whole frequency band of operation. The comparison of 2D radiation patterns obtained from HFSS and CST at 92, 94, and 96 GHz in both vertical plane (φ=90°) and horizontal plane (φ=0°) plane is shown in Figures 5 and 6, respectively. By increasing the ground plane size the small ripples in E-plane radiation pattern can be reduced, which are mainly due to diffraction of the limited ground plane. The cross polarization ratio of less than −20 dB is achieved in E- and H-plane radiation patterns, respectively. The antenna 3 dB beam width is found to be 50° and 45° in E- and H-plane, respectively.Figure 5
2D radiation pattern in H-plane with co- and cross polarization.
(a)
92 GHz
(b)
94 GHz
(c)
96 GHzFigure 6
2D radiation pattern in E-plane with co- and cross polarization.
(a)
92 GHz
(b)
94 GHz
(c)
96 GHzFrom the above results, it is noted that the peak gain obtained is only 7.5 dBi which is not enough for the point to point communication at 94 GHz. So we added a superstrate layer to improve the gain.


## 2.1. Design and Configuration

The single element SIW based structure of the proposed membrane antenna with 6 layers is shown in Figure1. The two substrates are utilized in design, that is, pyralux substrate with εr=2.4 and loss tangent tan⁡δ=0.002 and FR4 substrates with εr=4.4 and loss tangent tan⁡δ=0.02. The losses are incorporated in simulation. As shown in Figure 1(a) the top layer consists of a microstrip patch antenna (MPA) etched below the pyralux substrate. The second layer consists of the FR-4 substrate with an air cavity to support the top substrate layer. An air gap is drilled on the second layer to enhance the bandwidth of the antenna. We used the FR4 substrate because it is very cheap, low in loss, and compatible with mass production PCB technology electronics. The bottom three layers makes up a SIW slot antenna, where rectangular slot [16] in the top metallic layer of SIW is utilized to excite the MPA through the air gap made within the FR4 layer. The metallic via holes are made with in the bottom layer of pyralux substrate to form the SIW structure. The proposed membrane antenna structure in XZ plane is shown in Figure 1(b) for a better understanding of various layers. All copper layers used in proposed antenna structure have thickness of 18 μm, while the thickness of pyralux and FR-4 is taken to be 50 μm and 100 μm, respectively.Figure 1
Proposed membrane antenna geometry.
(a)
3D layered model
(b)
2D side viewThe distance between two rows of metallic via holes along with the dielectric between them determines the cutoff frequency. The SIW design generally works inTEn0 mode and does not support propagation of TM modes. For the dielectric filled waveguide (DFW) with same cutoff frequency, the broad side dimension of waveguide, that is, ad, is found by [17](1)ad=aεr,where εr is the dielectric constant of substrate and dimensions of “a” are taken from the standard WR-10 waveguide (i.e., 2.54 mm). Once the dimension ad for DFW is known, we can use the following equation to find the separation distance, that is, as, between the via rows of SIW [18]:(2)as=ad+d20.95p,where d is the diameter of metallic via holes connecting the upper and lower metallic layers of the bottom pyralux substrate. The via diameter (d) is taken by d=λg/5 and pitch, that is, centre to centre distance between the via holes, is taken by p<2d [19].In Figure2 the top view of proposed membrane antenna along with longitudinal slot placed on the top ground plane of SIW structure is shown. Usually the distance from the short circuited end of SIW to the centre of slot, that is, doff⁡2, is a quarter or odd multiple of quarter of the guided wavelength. Initially, this distance is chosen to be three quarter of the guided. Furthermore, the slot offset, that is, doff⁡1, is optimized for proper excitation of longitudinal slot as well as the metallic patch antenna on the top layer. The initial dimension for longitudinal slot length is taken by [17](3)Ls=λo2εr+1.Figure 2
Top view of SIW antenna along with rectangular slot.The pyralux substrate with patch on top layer is supported by a layer of FR4 substrate having an optimized air gap of 1.8 mm × 1.8 mm under the MPA. The optimized dimensions for MPA are found to be 1.12 × 0.99 mm. The total size of antenna is found to be 10 mm × 10 mm. A conventional 50 Ω microstrip line is utilized as a feed element in the proposed antenna. The microstrip to SIW transition consisting of a tapered microstrip is also optimized for proper impedance matching. The detailed view of proposed single membrane antenna along with various important design parameters for MPA and air gap in FR-4 substrate are well explained in [20] and in Figure 3.Figure 3
Top view of proposed membrane antenna geometry.The optimum dimension for the proposed membrane antenna structure isWo=0.145 mm, Wt=1.34 mm, Lo=1 mm, Lt=0.9 mm, d=0.5 mm, p=0.7 mm, as=1.5 mm, Ws=0.2 mm, Ls=1.225 mm, L=3.5 mm, doff⁡1=0.55 mm, doff⁡2=1.4125 mm, La=1.85 mm, Wa=1.85 mm, Lp=1.15 mm, Wp=1.05 mm, L=10 mm, and W=10 mm.


## 2.2. Results and Discussion

The proposed antenna structure is simulated and optimized using ANSYS High Frequency Structure Simulator (HFSS©). The results are further verified by simulating the proposed antenna structure in CST Microwave Studio.A comparison between the simulation results ofS11 and gain obtained from HFSS and CST are given in Figure 4. The two resonances at 92.4 GHz and 95.5 GHz are due to presence of slot and patch in antenna geometry, respectively, which are kept in close proximity to achieve wide impedance bandwidth. The FR-4 support with air gap plays an important role in merging the two resonances to achieve a wide impedance bandwidth. The antenna impedance bandwidth is found to be 5.4 GHz (91.5–96.9 GHz) from HFSS while from CST the impedance bandwidth is found to be approximately 4.9 GHz (91.5–96.4 GHz). The gain is found to be 7 dBi at the center frequency of 94 GHz. The peak gain of 7.5 dB is found at 95.5 GHz. Furthermore, the antenna gain remains above 6.7 dB in whole frequency band of operation.Figure 4
Simulated return lossS11 and realized gain.The antenna radiation pattern in both E-plane (horizontal plane) and H-plane (vertical plane) remains similar throughout the whole frequency band of operation. The comparison of 2D radiation patterns obtained from HFSS and CST at 92, 94, and 96 GHz in both vertical plane (φ=90°) and horizontal plane (φ=0°) plane is shown in Figures 5 and 6, respectively. By increasing the ground plane size the small ripples in E-plane radiation pattern can be reduced, which are mainly due to diffraction of the limited ground plane. The cross polarization ratio of less than −20 dB is achieved in E- and H-plane radiation patterns, respectively. The antenna 3 dB beam width is found to be 50° and 45° in E- and H-plane, respectively.Figure 5
2D radiation pattern in H-plane with co- and cross polarization.
(a)
92 GHz
(b)
94 GHz
(c)
96 GHzFigure 6
2D radiation pattern in E-plane with co- and cross polarization.
(a)
92 GHz
(b)
94 GHz
(c)
96 GHzFrom the above results, it is noted that the peak gain obtained is only 7.5 dBi which is not enough for the point to point communication at 94 GHz. So we added a superstrate layer to improve the gain.


## 3. Improvement of Gain with Superstrate Layer

Side view of proposed antenna with superstrate layer is shown in Figure7. It consists of the base antenna as explained in Section 2 and a dielectric superstrate layer is added above it. The material used for the superstrate is Roger RT-duroid 6010ML (εr=10.2, and tan⁡δ=0.003). A Rohacell foam layer of permittivity 1.05 is sandwiched between base antenna and superstrate as a support. The thickness t, dimension S, and the height of the superstrate h are studied and optimised as explained below.Figure 7
Cutting plane of a membrane antenna with superstrate layer, ground plane size10×10 mm2.The variation ofS11 and gain with different superstrate thickness (t = 0.127, 0.254, and 0.635 mm) with a constant height of h=0.55λ0 and size S=3.125×3.125λ0=10×10 mm2 is shown in Figure 8. It is noted that even though the maximum gain is achieved for a superstrate thickness of t=0.635 as compared to t=0.127λ0, the side lobe level (SLL) is worst as investigated in Figure 9. This poor SLL with large thickness of the material may be due to the surface waves at this high frequency. Hence we took the commercially available thickness t=0.127 mm as the superstrate material. The side lobe levels in this case are less than −15 dB in the whole frequency band of interest.Figure 8
Simulated results ofS11 and gain with various superstrate thickness for a fixed height h=0.55λ0, and size of the superstrate S=3.125×3.125λ0.Figure 9
The variation of side lobe level of the antenna with the variation of the thickness of the superstrate.Figure10 shows the variation of return loss (S11) and gain for different superstrate heights, h=0.5λ0, 0.55λ0, 0.6λ0, respectively, with a superstrate size of S=3.125×3.125λ0. It is clear from the figure that the maximum gain with worst bandwidth is achieved for a superstrate height of h=0.55λ0, while for the good desired bandwidth of 92 GHz–96 GHz is achieved with a height of h=0.6λ0 and is the best one. The high gain at h=0.55λ0 is because of the narrow bandwidth. We took the height h=0.6λ0 for the next optimisation simulation.Figure 10
Variation ofS11 and gain with a superstrate dimension of =3.125×3.125λ0, and with different heights of h=0.5λ0, h=0.55λ0, and h=0.6λ0.The variation of the superstrate sizeS for an optimised height of h=0.6λ0 is studied as shown in Figure 11. It is clear that the maximum gain of 14.4 dBi with good bandwidth of 92 GHz–96 GHz is achieved for a superstrate size of S=2λ0×2λ0.Figure 11
Variation ofS11 and gain for different dimensions size S with a fixed superstrate height of h=0.6λ0.The comparison of CST and HFSS results of return loss (S11), gain, and directivity of the optimised dimension is given in Figure 12. The 2 : 1 VSWR bandwidth is noted to be 92 GHz–96 GHz (i.e., 4.25% impedance bandwidth) with a maximum gain of 14.4 dBi and a directivity of 15.5 dBi at 94 GHz, hence a radiation efficiency of 77.6%. The results are in a good agreement between the CST and HFSS simulations. There is a gain enhancement of 7.2 dBi with the loaded superstrate layer as compared to without superstrate.Figure 12
Results ofS11, gain, and directivity over frequency with an optimised superstrate dimension S=2×2λ0, an optimised height h=0.6λ0, and an optimised thickness t=0.127 mm.The E-plane and H-plane radiation characteristics by using CST and HFSS simulations are shown in Figures13(a) and 13(b), respectively. The simulated results for the frequencies at 92 GHz, 94 GHz, and 96 GHz, respectively, are presented. Both simulation results are in good agreement. E-plane has the side lobe of level −14.8 dB and 3 dB beam width of 27.9° at 94 GHz. The H-plane has the side lobe of level −22.1 dB and 3 dB beam width of 25.1° at the same frequency. There is a cross polar level of less than −25 dB at all frequencies in both planes.Figure 13
Simulated (a) E-plane and (b) H-plane radiation characteristics of superstrate antenna for an optimised superstrate dimensionS=2λ0×2λ0, height h=0.6λ0, and thickness t=0.127 mm.
(a)
(b)


## 4. Conclusion

The design and the results of a superstrate loaded membrane antenna based on SIW technology are presented. A microstrip patch antenna along with a longitudinal slot is utilized to achieve a wide impedance bandwidth of 5 GHz at centre frequency of 94 GHz. The gain of the antenna for the single element is found to be above 6.5 dB in whole frequency band of operation and is further improved to a maximum of 14.4 dBi with an addition of superstrate layer. Furthermore, similar antenna radiation patterns are achieved in whole frequency band of operation. The proposed antenna finds applications in 94 GHz point-point communication systems.



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*Source: 385125-2015-10-01.xml*