# Three-Level DC-DC Controlled-Source Circuit of Marine Electromagnetic Detection Transmitter

Authors: Haijun Tao; Guopeng Zhang; Zheng Zheng
Journal: Journal of Electrical and Computer Engineering
 (2019)

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## Abstract

Nowadays, marine electromagnetic method is a geophysical method which can effectively explore natural gas hydrate (NGH) resources. This method acquires seafloor structure and distribution laws of mineral resources by transmitting high-power electromagnetic waves to the seafloor, and its detection depth relies on the intensity of transmitted electromagnetic waves. At present, marine electromagnetic transmitter is not proper for transmission of high-power electromagnetic waves, and therefore, a novel controlled-source circuit of soft switching three-level marine electromagnetic transmitter is proposed in this paper, one step-down winding is added at the secondary side of high-frequency transformer in series connection with a linear inductor, asymmetric phase-shift PWM control is used, and soft switching of controlled-source circuit within the overall power range is realized. A detailed analysis of the operation process and circuit characteristics is given, and finally, a prototype is designed. Experimental results indicate that this circuit is of wide ZVS range and high efficiency, and it meets requirements for deep-sea detection of high-voltage power supply transmitter.


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## Body

## 1. Introduction

During the exploratory development process of marine resources, due to high drilling cost and high investment risk, global major oil companies will carry out comprehensive prospecting work through multiple marine geophysical methods like an earthquake, gravity, and magnetic force before offshore drilling in order to reduce drilling risks of deepwater oil and gas resources [1]. The operation principle of marine controlled-source electromagnetic detection (MCSEM) is that electromagnetic transmitter is towed to offshore bottom and transmits high-power electromagnetic waves through transmitting dipole, field source signals are collected back by electromagnetic receiving equipment, and through signal processing and inversion, electro-conductive structural features of media below seafloor can be analyzed; it is one of the methods which can effectively explore natural gas hydrate (NGH) resources [2]. Due to attenuation of electromagnetic waves by seawater, the greater the current signals transmitted by the electromagnetic transmitter, the greater the detectable depth, and elevation of transmitting power will bring about problems of heat dissipation and energy loss in seafloor electromagnetic transmission system [3]. It is necessary to reduce internal resistance, reduce internal loss, and improve transmitting efficiency of the transmitter, and research and development of high-efficiency and large-current electromagnetic transmitter system are of great significance to deep-sea detection.The power supply of marine electromagnetic detection transmitter is a diesel generator which transmits electric energy to the underwater towed vehicle through kilometers of submarine cable transmission, and transmission voltage must be elevated in order to reduce the loss on transmission cable and match load impedance [4]. At present, high-power marine electromagnetic transmitter mainly adopts power frequency transformers + uncontrolled rectification or phase-controlled rectification modes with large volume, poor dynamic characteristic, and restricted transmission power [5]. The literature [6, 7] proposes using diode rectification + PWM full-bridge converter mode, which improves density and dynamic characteristic of transmitter power, but as transmitting power increases, rated voltage and rated current of switching devices also increase, which reduces switching frequency, enlarges transformer and filter volumes, and degrades power conversion efficiency. Multilevel technology can reduce voltage stress of switching devices; the study in [8] introduces cascading multilevel technology, but there exists a capacitor voltage balance problem. Pinheiro puts forward zero-voltage half-bridge three-level (HB-TL) converter, which reduces voltage stress of switching the device into half of the input voltage [9], and study [10] introduces half-bridge voltage balance control, but it is not appropriate for the high-power situation. Studies [11, 12] introduce hybrid-type full-bridge three-level converters, which reduces input and output current ripples and expands ZVS range of switching devices, as this circuit includes three-level bridge arm and two-level bridge arm, and voltage stress of switching device with two-level bridge arm is input voltage, it is inapplicable to high-voltage input. Literature [13, 14] propose a full-bridge three-level (FB-TL) converter which consists of 8 switching devices, voltage stress of each switching device is half of the input voltage, but the lagging leg is of difficult current commutation and low efficiency. An improved full-bridge three-level converter is introduced and applied to the controlled-source circuit of high-power marine electromagnetic detection transmitter. High-frequency transformer has three secondary windings, where two step-down windings do rectification in parallel so that output current is elevated, another step-down winding is connected to commutation inductor, and this solves difficult current commutation problems of lagging leg under no-load or light-load conditions, realizes soft switching of controlled-source circuit within overall power range, and reduces volumes of input and output filters.


## 2. Operation Principle of Marine Electromagnetic Detection Transmitter

Marine controlled-source electromagnetic detection system mainly consists of two parts—transmitter and receiver, the former includes diesel generator, three-phase to single-phase power supplies as well as underwater electromagnetic transmitter, and dipolar dragging system, while the latter is a seafloor electromagnetic signal receiving station as shown in Figure1. The specific operation process is firstly electromagnetic acquisition station is placed according to preset coordinate points, the automatic recording is started, and then the ship will drag underwater electromagnetic transmitter to do the movement at a uniform speed of 3∼5 knots according to preset track. Dipole connected to an electromagnetic transmitter will continuously excite electromagnetic waves with different frequencies, and the ship will make a circulation according to the track in order to improve the signal-to-noise ratio of acquired data. Signals will be sent to electromagnetic signal acquisition station which will then automatically release counterweight and emerge from the water after completion of data acquisition. Exploration of the next array or survey line will be implemented.Figure 1
Schematic diagram of operation method of marine controlled-source electromagnetic exploration.Marine electromagnetic transmitter, a key component of marine controlled-source electromagnetic exploration system, is responsible for transmitting high-power electromagnetic waves to the seafloor, and it is mainly divided into two parts, namely, shipboard part and underwater part.


### 2.1. Shipboard Part

As shown in Figure2, diesel generator supplies power, and inversion is implemented to obtain single-phase sine alternating current after three-phase rectification and filtering and provide high-voltage single-phase alternating current for underwater tow after the boost transformer T2.Figure 2
Shipboard power supply circuit of marine transmitter.


### 2.2. Underwater Part

Underwater part mainly consists of the controlled-source circuit, transmitting bridge, and transmitting dipole, where the controlled-source circuit is responsible for the transformation of electric energy, namely, transforming single-phase alternating current input by tow into controlled direct current, and transmitting bridge generates time-domain or frequency-domain electromagnetic wave which will be excited by transmitting dipole to the seafloor. At present, the controlled-source circuit of marine electromagnetic transmitter is diode uncontrolled or thyristor phase-controlled rectification type as shown in Figure3.Figure 3
Controlled-source transmitter circuit using phase-controlled rectification.Shipboard single-phase power supply is transmitted to underwater tow through photoelectric composite cable, single-phase alternating current is regulated by the power frequency step-down transformer inside the tow, diode uncontrolled rectification or thyristor phase-controlled rectification and filtering are used to obtain low-voltage direct current, and then frequency-controllable square wave alternating current will be generated through inversion of transmitting bridge and excited to the seafloor through transmitting dipole. Diode uncontrolled rectification technology is quite simple and has been widely applied in the high-power transmitter. As uncontrolled rectification circuit cannot regulate voltage amplitude, it usually controls the output voltage through generator excitation, but excitation regulation will influence the revolving speed of the generator, which results in large voltage fluctuation. Consequently, dynamic response of regulation is slow, and thyristors can replace power diodes to constitute phase-controlled rectifier bridge which can control voltage amplitude, but it will cause reduction of power factor.Therefore, the PWM converter is proposed to replace uncontrolled or phase-controlled rectifier as shown in Figure4. Single-phase alternating current is inverted into high-frequency alternating current after diode rectification through the high-frequency isolation step-down transformer, it is rectified into low-voltage direct current, and then a kind of time-domain or frequency-domain electromagnetic waves are output through the transmitting bridge. When input voltage or load current goes through disturbance, the circuit output will be influenced, so it is necessary to design a closed-loop control system to regulate the width of output square wave impulse voltage of inverter bridge in order to maintain output stability so as to control the output voltage.Figure 4
Transmitter circuit using PWM converter.Compared with the controlled-source circuit of uncontrolled or phase-controlled rectification mode, PWM transformation mode seems quite complicated but with unique advantages: (1) functions of the high-frequency transformer are mainly wide-range voltage regulation realized separately, its operating frequency is tens of kHz or even higher, and transformer volume is greatly reduced; (2) high control precision: control cycle of the phase-controlled rectification-type circuit is half of single-phase AC cycle, while control cycle of PWM transformation-type circuit is a switching cycle, so the difference between the two reaches dozens of times, and the control performance of PWM transformation-type circuit is obviously improved; (3) as operation frequency is high, filter volume is also greatly reduced so that the power density of the transmitter is improved.High-power PWM transformer is generally full-bridge (ZVS-FB) PWM converter, and in this case, the capacity of switching device must be increased in order to improve transmitting power, which will then cause reduction of switching frequency, enlargement of transformer and filter volumes, and reduction of transformation efficiency of electric energy, so it cannot meet deep marine exploration requirements. Then a full-bridge three-level (TL) DC/DC controlled-source circuit will be introduced and switching device bears 1/2 of input voltage, and meanwhile, switching device can realize soft switching within the whole power range to reduce switching loss and improve circuit efficiency. The circuit can meet the requirements of high-power electromagnetic wave transmission with input voltage 1500 V and output current 500 A.


## 2.1. Shipboard Part

As shown in Figure2, diesel generator supplies power, and inversion is implemented to obtain single-phase sine alternating current after three-phase rectification and filtering and provide high-voltage single-phase alternating current for underwater tow after the boost transformer T2.Figure 2
Shipboard power supply circuit of marine transmitter.


## 2.2. Underwater Part

Underwater part mainly consists of the controlled-source circuit, transmitting bridge, and transmitting dipole, where the controlled-source circuit is responsible for the transformation of electric energy, namely, transforming single-phase alternating current input by tow into controlled direct current, and transmitting bridge generates time-domain or frequency-domain electromagnetic wave which will be excited by transmitting dipole to the seafloor. At present, the controlled-source circuit of marine electromagnetic transmitter is diode uncontrolled or thyristor phase-controlled rectification type as shown in Figure3.Figure 3
Controlled-source transmitter circuit using phase-controlled rectification.Shipboard single-phase power supply is transmitted to underwater tow through photoelectric composite cable, single-phase alternating current is regulated by the power frequency step-down transformer inside the tow, diode uncontrolled rectification or thyristor phase-controlled rectification and filtering are used to obtain low-voltage direct current, and then frequency-controllable square wave alternating current will be generated through inversion of transmitting bridge and excited to the seafloor through transmitting dipole. Diode uncontrolled rectification technology is quite simple and has been widely applied in the high-power transmitter. As uncontrolled rectification circuit cannot regulate voltage amplitude, it usually controls the output voltage through generator excitation, but excitation regulation will influence the revolving speed of the generator, which results in large voltage fluctuation. Consequently, dynamic response of regulation is slow, and thyristors can replace power diodes to constitute phase-controlled rectifier bridge which can control voltage amplitude, but it will cause reduction of power factor.Therefore, the PWM converter is proposed to replace uncontrolled or phase-controlled rectifier as shown in Figure4. Single-phase alternating current is inverted into high-frequency alternating current after diode rectification through the high-frequency isolation step-down transformer, it is rectified into low-voltage direct current, and then a kind of time-domain or frequency-domain electromagnetic waves are output through the transmitting bridge. When input voltage or load current goes through disturbance, the circuit output will be influenced, so it is necessary to design a closed-loop control system to regulate the width of output square wave impulse voltage of inverter bridge in order to maintain output stability so as to control the output voltage.Figure 4
Transmitter circuit using PWM converter.Compared with the controlled-source circuit of uncontrolled or phase-controlled rectification mode, PWM transformation mode seems quite complicated but with unique advantages: (1) functions of the high-frequency transformer are mainly wide-range voltage regulation realized separately, its operating frequency is tens of kHz or even higher, and transformer volume is greatly reduced; (2) high control precision: control cycle of the phase-controlled rectification-type circuit is half of single-phase AC cycle, while control cycle of PWM transformation-type circuit is a switching cycle, so the difference between the two reaches dozens of times, and the control performance of PWM transformation-type circuit is obviously improved; (3) as operation frequency is high, filter volume is also greatly reduced so that the power density of the transmitter is improved.High-power PWM transformer is generally full-bridge (ZVS-FB) PWM converter, and in this case, the capacity of switching device must be increased in order to improve transmitting power, which will then cause reduction of switching frequency, enlargement of transformer and filter volumes, and reduction of transformation efficiency of electric energy, so it cannot meet deep marine exploration requirements. Then a full-bridge three-level (TL) DC/DC controlled-source circuit will be introduced and switching device bears 1/2 of input voltage, and meanwhile, switching device can realize soft switching within the whole power range to reduce switching loss and improve circuit efficiency. The circuit can meet the requirements of high-power electromagnetic wave transmission with input voltage 1500 V and output current 500 A.


## 3. Three-Level Controlled-Source Circuit

### 3.1. Circuit Topology

A full-bridge three-level (FB-TL) converter is introduced as shown in Figure5 in reference [15], where switching devices S1∼S4 (including D1∼D4 and Cs1∼Cs4), clamping diodes D9 and D10, and flying capacitor Css1 constitute left-side bridge arm, switching devices S5∼S8 (including D5∼D8 and Cs5∼Cs8), clamping diodes D11 and D12, and flying capacitor Css2 constitute right-side bridge arm and two bridge arms jointly input voltage-divider capacitors Cd1 and Cd2, Ls is the resonant inductor, n is the turn ratio of transformer primary winding to rectification winding. The circuit is suitable for high-voltage and high-power applications since each switch is subjected to half of the input voltage.Figure 5
The conventional full-bridge three-level circuit.However, the circuit with multiple lagging legs is difficult to commutate. If the resonant inductanceLs is increased directly, serious duty cycle loss and circulating current loss will occur. Therefore, an improved full-bridge three-level circuit is proposed as a DC/DC controllable source circuit as shown in Figure 6. It has two step-down windings rectifiers in parallel, and one step-down winding was added at the secondary side of the high-frequency transformer in series connection with a linear inductor Lc, and Lr is the leakage inductor of the high-frequency transformer. The former is used to increase the output current; the latter is used to ensure the devices soft switching in full power scope. The auxiliary inductor Lc at a secondary winding can reduce the loss on commutation inductor due to the low voltage and current at a secondary winding and reduce the influence of duty cycle loss.Figure 6
The improved three-level controlled-source circuit.


### 3.2. Analysis of the Operation Process

During the operation process analysis, it is assumed that(1)
All power diodes, switching devices, inductors, and capacitors are ideal devices(2)
Cs1=Cs4=Cchop, Cs2=Cs3=Cs5=Cs6=Cs7=Cs8=Clag, Css1=Css2=Css, Css≫Cchop, Css≫Clag(3)
L≫Lr/n2, where L is filter inductance, and it is large enough and can be regarded as a constant current source(4)
Parameters of two rectifying-circuit devices at the secondary side of the transformer are consistent under the same operation mode, and the only single rectifying circuit is taken into consideration in the analysisThree conditions should be met by the optimal control mode: (1) power transmission is the maximum under the same duty ratio; (2) filtering inductive current pulsation is the minimum; and (3) switching devices realize soft switching [15] under asymmetric phase-shift PWM control mode, and main operation waveforms are as shown in Figure 7. Turn-on times of S1 and S2, S7, and S8 (or S4 and S3, S5, and S6) are the same but with different turn-off times. PWM (pulse width modulation) is used for drive signals of S1 S4, while signals driven by lagging leg maintain maximum pulse width. The output voltage is controlled by modulating pulse widths of drive signals S1 and S4. S1 and S4 are called leading legs, and other switching devices are lagging legs. This control mode is easy for the realization of digitalization, which overcomes problems like control precision and poor flexibility of dedicated chips for traditional phase-shift control. This circuit totally has 14 operation modes.Figure 7
Operation waveforms of three-level controlled-source circuit.Mode 0.
t
0
− (Figure 8).
Att0−, switching devices S1, S2, S7, and S8 are turned on, the primary current of the transformer is ip=I0/n, the voltage at AB points is uAB=Ui, rectifying diode DR1 is turned on, and DR1 is turned off. The voltage at two ends of capacitors Cd1 and Cd2 in parallel is Ui/2, where Cd1 charges flying capacitor Css1 through branches S1, Css1, and D10 and Cd2 charges flying capacitor Css2 through branches S8, Css2, and D11 until Ui/2. Under the function of flying capacitors, voltage stresses of S3, S4, S5, and S6 are all Ui/2. Commutation inductor current starts being increased in a forward direction from maximum reverse current ILc0, and commutation inductor current is as follows:(1)iLct=−ILc0+1Lc∫0t0Uimdt,where m is turn ratio of transformer primary winding to commutation winding.
As the output filter inductanceL is large enough, the output current Io is approximately constant within one switching period. Therefore, the primary current ip of transformer consists of conversion value of commutation inductor current iLc, conversion value of output current Io, and transformer leakage-inductance current iLr, namely,(2)ipt=Ion+iLctm+iLrt.Figure 8
Equivalent circuit of mode 0.Mode 1.
[t0+, t1] (Figure 9).
Att0, the switching device S1 is turned off, and primary current ip is transferred from S1 to its shunt capacitor Cs1 to charge Cs1. As the flying capacitor Css1 is in parallel with S2 and S3, voltage sum of Cs1 and Cs4 is Ui/2, and Cs4 is discharged when Cs1 is charged. As Cs1 starts increasing from zero, S1 is turned off at zero voltage. In this stage, the primary side outputs power to the secondary side, as filter inductance is large enough, the primary current ip of the transformer basically remains constant, and Cs1 voltage increases linearly while Cs4 voltage decreases linearly.(3)uCs1t=Ip02Cleadt−t0,uCs4t=Ui2−uCs1=Ui2−Ip02Cleadt−t0.
Tillt1, the voltage of Cs4 declines to zero and the voltage of Cs1 is charged to Ui/2. In this case, the commutation inductor current is(4)iLct=iLct0+1Lc∫t0t1Ui/2+uCs4mdt.Figure 9
Equivalent circuit of mode 1.Mode 2.
[t1, t2] (Figure 10).
Att1, as Cs4 voltage reduces to zero, D4 is naturally turned on. If the voltage of the flying capacitor Css1 is slightly reduced, then D9 is turned on, and Cs4 provides energy for the load. In this stage, under the function of flying capacitors, Cs4 voltage maintains zero, so S4 can be turned on at any time before t5 at zero voltage, which indicates that asymmetric phase-shift PWM control is feasible. In this case, the voltage at AB points is Ui/2, and commutation inductor current is(5)iLct=iLct1+1Lc∫t1t2Ui2mdt.Figure 10
Equivalent circuit of mode 2.Mode 3.
[t2, t3] (Figure 11).
Att2, S2, S7, and S8 are turned off. At left-side bridge arm, ip is transferred from S2 to charge Cs2. Under the function of flying capacitors, Cs3 will certainly be discharged while Cs2 is charged, and S2 is turned off at zero voltage. At right-side bridge arm, ip is transferred from S7 and S8 to Cs5, Cs6, Cs7, and Cs8 to charge Cs7 and Cs8 to discharge Cs5 and Cs6, and S7 S8 are turned off at zero voltage. Up to t3, uAB declines to zero. In this stage, the primary side still outputs power to the secondary side, the primary current ip is basically constant, so both charging and discharging processes of capacitors are linear.(6)uCs2t=Ip02Clag∗t−t2,uCs3t=Ui2−uCs2t=Ui2−Ip02Clag∗t−t2,uCs7t=uCs8t=Ip02Clag∗t−t2,uCs5t=uCs6t=Ui2−Ip02Clag∗t−t2.
At the time, converting inductor current is(7)iLct=iLct2+1Lc∫t2t3Ui/2−uCs2mdt.
Up tot3, uAB declines to zero, and commutation inductor current reaches peak value ILc0. Here, uCs3=Ui/3, uCs2=Ui/6, uCs7=uCs8=Ui/6, and uCs5=uCs6=Ui/3.Figure 11
Equivalent circuit of mode 3.Mode 4.
[t3, t4] (Figure 12).
Att3, uAB declines to zero, so does secondary winding voltage, and rectifying diodes are all turned on. As at the time commutation inductor current reaches peak value, commutation inductor current charges Cs2, Cs7, and Cs8 discharges Cs3, Cs5, and Cs6 until t4 when voltages of Cs3, Cs5, and Cs6 are reduced to zero. Commutation inductor and a number of windings are reasonably designed, so soft switching of S2, S7, and S8 can be easily realized, and then(8)ipt=Ip0cosω1t−t3,uCs2t=Ui6+Ip02∗2Lr+Lc3Clagsinω1t−t3,uCs3t=Ui3−Ip02∗2Lr+Lc3Clagsinω1t−t3,uCs7t=uCs8t=Ui6+Ip02∗2Lr+Lc3Clagsinω1t−t3,uCs5t=uCs6t=Ui3−Ip02∗2Lr+Lc3Clagsinω1t−t3,where ω1=1/2Lr+LcClag/2.
Att4, the value ip is(9)Ipt4=Ip02−2Clag3Lr+LcUi2.Figure 12
Equivalent circuit of mode 4.Mode 5.
[t4, t5] (Figure 13).
Att4, currents of commutation inductor do not decline to zero yet; D3, D5, and D6 are naturally turned on, current ip continues to reduce, and up to t5, it presents linear decline to zero, so S3, S4, S5, and S6 can be turned on at zero voltage at any time in this stage. Current ip is(10)ip=Ipt4−UiLrt−t4−UimLct−t4.Figure 13
Equivalent circuit of mode 5.Mode 6.
[t5, t6] (Figure 14).
Att5, S3, S4, S5, and S6 are turned on, and ip starts being increased in the inverse direction. In this case, ip does not reach a primary converted value of load current Ip0, two rectifying diodes are still simultaneously turned on, ip and increases in the inverse direction. Up to t6, the absolute value of ip increases to Ip0.Figure 14
Equivalent circuit of mode 6.Mode 7.
[t6, t7] (Figure 15).
Att6, as ip reaches Ip0, rectifying diode DR1 is turned off, and the transformer provides energy for the load through DR2. As during the switching process, energies of Css1 and Css2 are somewhat released, and at the time, there will be a process in which Cd2 charges Css1 through branches D9, Css1, and S4. Cd1 charges Css2 through branches D12, Css2, and S5. Up to t7, the circuit starts the operation process of the next half period, which will not be described in details.Figure 15
Equivalent circuit of mode 7.


## 3.1. Circuit Topology

A full-bridge three-level (FB-TL) converter is introduced as shown in Figure5 in reference [15], where switching devices S1∼S4 (including D1∼D4 and Cs1∼Cs4), clamping diodes D9 and D10, and flying capacitor Css1 constitute left-side bridge arm, switching devices S5∼S8 (including D5∼D8 and Cs5∼Cs8), clamping diodes D11 and D12, and flying capacitor Css2 constitute right-side bridge arm and two bridge arms jointly input voltage-divider capacitors Cd1 and Cd2, Ls is the resonant inductor, n is the turn ratio of transformer primary winding to rectification winding. The circuit is suitable for high-voltage and high-power applications since each switch is subjected to half of the input voltage.Figure 5
The conventional full-bridge three-level circuit.However, the circuit with multiple lagging legs is difficult to commutate. If the resonant inductanceLs is increased directly, serious duty cycle loss and circulating current loss will occur. Therefore, an improved full-bridge three-level circuit is proposed as a DC/DC controllable source circuit as shown in Figure 6. It has two step-down windings rectifiers in parallel, and one step-down winding was added at the secondary side of the high-frequency transformer in series connection with a linear inductor Lc, and Lr is the leakage inductor of the high-frequency transformer. The former is used to increase the output current; the latter is used to ensure the devices soft switching in full power scope. The auxiliary inductor Lc at a secondary winding can reduce the loss on commutation inductor due to the low voltage and current at a secondary winding and reduce the influence of duty cycle loss.Figure 6
The improved three-level controlled-source circuit.


## 3.2. Analysis of the Operation Process

During the operation process analysis, it is assumed that(1)
All power diodes, switching devices, inductors, and capacitors are ideal devices(2)
Cs1=Cs4=Cchop, Cs2=Cs3=Cs5=Cs6=Cs7=Cs8=Clag, Css1=Css2=Css, Css≫Cchop, Css≫Clag(3)
L≫Lr/n2, where L is filter inductance, and it is large enough and can be regarded as a constant current source(4)
Parameters of two rectifying-circuit devices at the secondary side of the transformer are consistent under the same operation mode, and the only single rectifying circuit is taken into consideration in the analysisThree conditions should be met by the optimal control mode: (1) power transmission is the maximum under the same duty ratio; (2) filtering inductive current pulsation is the minimum; and (3) switching devices realize soft switching [15] under asymmetric phase-shift PWM control mode, and main operation waveforms are as shown in Figure 7. Turn-on times of S1 and S2, S7, and S8 (or S4 and S3, S5, and S6) are the same but with different turn-off times. PWM (pulse width modulation) is used for drive signals of S1 S4, while signals driven by lagging leg maintain maximum pulse width. The output voltage is controlled by modulating pulse widths of drive signals S1 and S4. S1 and S4 are called leading legs, and other switching devices are lagging legs. This control mode is easy for the realization of digitalization, which overcomes problems like control precision and poor flexibility of dedicated chips for traditional phase-shift control. This circuit totally has 14 operation modes.Figure 7
Operation waveforms of three-level controlled-source circuit.Mode 0.
t
0
− (Figure 8).
Att0−, switching devices S1, S2, S7, and S8 are turned on, the primary current of the transformer is ip=I0/n, the voltage at AB points is uAB=Ui, rectifying diode DR1 is turned on, and DR1 is turned off. The voltage at two ends of capacitors Cd1 and Cd2 in parallel is Ui/2, where Cd1 charges flying capacitor Css1 through branches S1, Css1, and D10 and Cd2 charges flying capacitor Css2 through branches S8, Css2, and D11 until Ui/2. Under the function of flying capacitors, voltage stresses of S3, S4, S5, and S6 are all Ui/2. Commutation inductor current starts being increased in a forward direction from maximum reverse current ILc0, and commutation inductor current is as follows:(1)iLct=−ILc0+1Lc∫0t0Uimdt,where m is turn ratio of transformer primary winding to commutation winding.
As the output filter inductanceL is large enough, the output current Io is approximately constant within one switching period. Therefore, the primary current ip of transformer consists of conversion value of commutation inductor current iLc, conversion value of output current Io, and transformer leakage-inductance current iLr, namely,(2)ipt=Ion+iLctm+iLrt.Figure 8
Equivalent circuit of mode 0.Mode 1.
[t0+, t1] (Figure 9).
Att0, the switching device S1 is turned off, and primary current ip is transferred from S1 to its shunt capacitor Cs1 to charge Cs1. As the flying capacitor Css1 is in parallel with S2 and S3, voltage sum of Cs1 and Cs4 is Ui/2, and Cs4 is discharged when Cs1 is charged. As Cs1 starts increasing from zero, S1 is turned off at zero voltage. In this stage, the primary side outputs power to the secondary side, as filter inductance is large enough, the primary current ip of the transformer basically remains constant, and Cs1 voltage increases linearly while Cs4 voltage decreases linearly.(3)uCs1t=Ip02Cleadt−t0,uCs4t=Ui2−uCs1=Ui2−Ip02Cleadt−t0.
Tillt1, the voltage of Cs4 declines to zero and the voltage of Cs1 is charged to Ui/2. In this case, the commutation inductor current is(4)iLct=iLct0+1Lc∫t0t1Ui/2+uCs4mdt.Figure 9
Equivalent circuit of mode 1.Mode 2.
[t1, t2] (Figure 10).
Att1, as Cs4 voltage reduces to zero, D4 is naturally turned on. If the voltage of the flying capacitor Css1 is slightly reduced, then D9 is turned on, and Cs4 provides energy for the load. In this stage, under the function of flying capacitors, Cs4 voltage maintains zero, so S4 can be turned on at any time before t5 at zero voltage, which indicates that asymmetric phase-shift PWM control is feasible. In this case, the voltage at AB points is Ui/2, and commutation inductor current is(5)iLct=iLct1+1Lc∫t1t2Ui2mdt.Figure 10
Equivalent circuit of mode 2.Mode 3.
[t2, t3] (Figure 11).
Att2, S2, S7, and S8 are turned off. At left-side bridge arm, ip is transferred from S2 to charge Cs2. Under the function of flying capacitors, Cs3 will certainly be discharged while Cs2 is charged, and S2 is turned off at zero voltage. At right-side bridge arm, ip is transferred from S7 and S8 to Cs5, Cs6, Cs7, and Cs8 to charge Cs7 and Cs8 to discharge Cs5 and Cs6, and S7 S8 are turned off at zero voltage. Up to t3, uAB declines to zero. In this stage, the primary side still outputs power to the secondary side, the primary current ip is basically constant, so both charging and discharging processes of capacitors are linear.(6)uCs2t=Ip02Clag∗t−t2,uCs3t=Ui2−uCs2t=Ui2−Ip02Clag∗t−t2,uCs7t=uCs8t=Ip02Clag∗t−t2,uCs5t=uCs6t=Ui2−Ip02Clag∗t−t2.
At the time, converting inductor current is(7)iLct=iLct2+1Lc∫t2t3Ui/2−uCs2mdt.
Up tot3, uAB declines to zero, and commutation inductor current reaches peak value ILc0. Here, uCs3=Ui/3, uCs2=Ui/6, uCs7=uCs8=Ui/6, and uCs5=uCs6=Ui/3.Figure 11
Equivalent circuit of mode 3.Mode 4.
[t3, t4] (Figure 12).
Att3, uAB declines to zero, so does secondary winding voltage, and rectifying diodes are all turned on. As at the time commutation inductor current reaches peak value, commutation inductor current charges Cs2, Cs7, and Cs8 discharges Cs3, Cs5, and Cs6 until t4 when voltages of Cs3, Cs5, and Cs6 are reduced to zero. Commutation inductor and a number of windings are reasonably designed, so soft switching of S2, S7, and S8 can be easily realized, and then(8)ipt=Ip0cosω1t−t3,uCs2t=Ui6+Ip02∗2Lr+Lc3Clagsinω1t−t3,uCs3t=Ui3−Ip02∗2Lr+Lc3Clagsinω1t−t3,uCs7t=uCs8t=Ui6+Ip02∗2Lr+Lc3Clagsinω1t−t3,uCs5t=uCs6t=Ui3−Ip02∗2Lr+Lc3Clagsinω1t−t3,where ω1=1/2Lr+LcClag/2.
Att4, the value ip is(9)Ipt4=Ip02−2Clag3Lr+LcUi2.Figure 12
Equivalent circuit of mode 4.Mode 5.
[t4, t5] (Figure 13).
Att4, currents of commutation inductor do not decline to zero yet; D3, D5, and D6 are naturally turned on, current ip continues to reduce, and up to t5, it presents linear decline to zero, so S3, S4, S5, and S6 can be turned on at zero voltage at any time in this stage. Current ip is(10)ip=Ipt4−UiLrt−t4−UimLct−t4.Figure 13
Equivalent circuit of mode 5.Mode 6.
[t5, t6] (Figure 14).
Att5, S3, S4, S5, and S6 are turned on, and ip starts being increased in the inverse direction. In this case, ip does not reach a primary converted value of load current Ip0, two rectifying diodes are still simultaneously turned on, ip and increases in the inverse direction. Up to t6, the absolute value of ip increases to Ip0.Figure 14
Equivalent circuit of mode 6.Mode 7.
[t6, t7] (Figure 15).
Att6, as ip reaches Ip0, rectifying diode DR1 is turned off, and the transformer provides energy for the load through DR2. As during the switching process, energies of Css1 and Css2 are somewhat released, and at the time, there will be a process in which Cd2 charges Css1 through branches D9, Css1, and S4. Cd1 charges Css2 through branches D12, Css2, and S5. Up to t7, the circuit starts the operation process of the next half period, which will not be described in details.Figure 15
Equivalent circuit of mode 7.


## 4. Circuit Characteristics

### 4.1. Functions of Flying Capacitors and Flywheel Diodes

The operation process of full-bridge three-level DC/DC controlled-source circuit mentioned above has made a detailed analysis of flying capacitors and flywheel diodes at left-side bridge arm. Now functions of flying capacitors and flywheel diodes at right bridge arm will analyze. It can be known from driving waveforms in Figure7 that switching devices S5 and S6 (or S7 and S8) of right three-level bridge arm are simultaneously turned on and turned off; however, in the actual circuit, differences will certainly exist in turning on and turning off of S5 and S6 (or S7,S8). If there are no flying capacitors or flywheel diodes, it will possibly cause nonuniform voltage stress of switching devices. When switching device is turned on at ZVS, its antiparallel diodes have been turned on, and at the time, differences existing in turning on of switching devices will not cause nonuniformity of voltage stress of switching devices. Functions of flying capacitors and flywheel diodes will be mainly discussed under differences of turning off of switching devices.It is assumed that switching devicesS5 and S6 are simultaneously turned on and turned off; switching devices S7 and S8 will be simultaneously turned on but they cannot be simultaneously turned off. The discussion will be made under two circumstances: S7 is turned off before and after S8.


#### 4.1.1. Switching DeviceS7 Is Turned Off before S8

Figure16 gives main waveforms when switching device S7 is turned off before S8. Before t0, switching devices S1, S2, S7, and S8 are turned on. At t0, switching devices S7 and S8 should be simultaneously turned off at first; the switching device S7 is turned off while S8 is still under on state due to the difference of characteristics of circuit devices. ip charges capacitor Cs7 and discharges the capacitor Cs6 through a flying capacitor Css2 and ip reduces.Figure 16
Main waveforms whenS7 is turned off before S8.Att1, S8 is turned off, and ip charges the capacitor Cs8 and discharges the capacitor Cs5 through the flying capacitor Css2. Meanwhile, ip continues to charge the capacitor Cs7 and discharge Cs6 through the flying capacitor Css2.(11)ip=iCs6+iCs7,(12)iCs5=iCs6+iCss2,(13)iCs8=iCs7−iCss2.Through the addition of formula (12) and (13), the following can be obtained by combining (11):(14)ip=iCs5+iCs8=iCs6+iCs7.It can be known from formula (14) that the current which discharges shunt capacitor of the lagging leg is ip and energy which realizes ZVS of the lagging leg is provided by commutation inductor.Att2, the voltage of the capacitor Cs6 reduces to zero if the primary current does not reduce to zero yet; the diode D6 is naturally turned on, and the switching device S6 can be turned on at zero voltage. ip continues to charge the capacitor Cs8 through a flying capacitor Css2 and discharge capacitor Cs5. At t3, the voltage of the capacitor Cs5 reduces to zero, if the primary current does not reduce to zero yet, the diode D5 will be naturally turned on, and switching device S5 can be turned on at zero voltage. As driving signals of S5 and S6 are the same, turning-on signals are sent to S5 and S6 at t3, which can be turned on at zero voltage.


#### 4.1.2. Switching DeviceS7 Is Turned Off after S8

Figure17 gives main waveforms when switching device S7 is turned off after S8. Before t0, the switching device S8 is turned off while S7 is still turned on, primary current ip charges capacitor Cs7 and discharges the capacitor Cs5 through a flying capacitor Css2, and ip reduces. At t1, the switching device S7 is turned off, ip charges capacitor Cs7 and discharges the capacitor Cs6 through the flying capacitor Css2. Meanwhile, ip continues to charge Cs8 and discharge the capacitor Cs6 through the flying capacitor Css2. Formula (14) indicates that the current which discharges shunt capacitance of lagging leg is ip, and the energy which realizes ZVS of the lagging leg is provided by commutation inductor. At t2, the voltage of the capacitor Cs5 reduces to zero if the primary current does not decline to zero yet, the diode D5 is naturally turned on, and the switching device S5 can be turned on at zero voltage. ip continues to charge the capacitor Cs7 through a flying capacitor Css2 and discharge capacitor Cs7. At t3, the voltage of the capacitor Cs6 reduces to zero, the diode D6 is naturally turned on, and the switching device S6 can be turned on at zero voltage.Figure 17
Main waveforms whenS7 is turned off after S8.For right three-level bridge arm, if there is no flying capacitor or flywheel diode, voltage stress of the switching device which is turned off firstly will exceedUi/2 when the above two switching devices or below two switching devices are not simultaneously turned off, which will cause an imbalance of voltage stress borne by switching devices.The object of flywheel diode is to provide channel supplementing energy for flying capacitor and to make the voltage of the flying capacitor maintain atUi/2. It can be seen from the above analysis that flying capacitor not only realizes decoupling of charging and discharging processes of parallel capacitors of switching devices but also keeps voltage stress of switching devices at Ui/2, which avoids an imbalance of voltage stress caused by differences of turning on and turning off of two pairs of switching devices.


### 4.2. Conditions of Soft Switching

#### 4.2.1. Soft Switching of the Leading Leg

It can be known from the previous analysis that to realize ZVS, the leading leg must have enough energy to take away electric charge on a parallel capacitorCs4 of switching device S4 and discharge parallel capacitor Cs1 of switching device S1 which is turned off; then, the energy used by the leading leg to realize ZVS is(15)Echop≥12Cs1Ui22+12Cs4Ui22=14CchopUi2.This energy mainly comes from primary converted value from load current. As output filter inductance is large enough, it is approximately believed that the current converted from constant current source to the primary side will easily let leading leg be turned on at zero voltage.


#### 4.2.2. Soft Switching of Lagging Leg

Three-level DC/DC controlled-source circuit has three lagging devices which are simultaneously turned off. During the turning off process, ZVS of the lagging leg can be realized only by charging parallel capacitors on lagging device which is turned off and by completely discharging parallel capacitors on three lagging devices. Taking turning off ofS2, S7, and S8 as an example, they are divided into two stages as shown in Figure 3. In stage I, uAB reduces to zero, and corresponding to mode 3, energy is mainly provided by the current converted from output filter inductor in this stage:(16)Elag1≥12Cs2+Cs7+Cs8Udc32+12Cs3+Cs5+Cs6Udc62=524ClagUdc2.In Stage II,uAB rises to Ui, and corresponding to mode 4, energy is mainly provided by the current converted from commutation inductor in this stage.(17)Elag2≥12Cs2+Cs7+Cs8Udc22−Udc62+12Cs3+Cs5+Cs6Udc32=12ClagUdc2.The energy of the commutation inductor must satisfy the following expression:(18)12LcUdcTs4mLc2=12ClagUdc2.Then(19)Lc=Ts216m2Clag.If ZVS is realized when the output current is 1/5 of the designed full load, the duty cycle loss of conventional three-level converter is as follows:(20)Lc′=25Clagn2Udc2Io2.


### 4.3. Duty Ratio Loss

Due to the existence of leakage inductance of the transformer, it will certainly bring about duty ratio loss. It can be known from previous analysis that inverted output voltageuAB has reached −Ui at t4, and this time is corresponding controlled duty ratio, but voltage us of secondary side of transformer does not rise to nUdc until t6, and this time is corresponding to effective duty ratio, and difference value between the two times is time of duty ratio loss as shown in Figure 3.(21)t45=Ipt4LrUi,t56=Ip0LrUi.Duty ratio loss is as follows:(22)Dloss=t45+t56Ts=2Ipt4+Ip0LrTsUi.The duty cycle loss of conventional three-level converter is as follows:(23)Dloss′=2Ipt4+Ip0LsTsUi.As commutation inductor is used to provide energy for turning off of lagging leg, leakage inductance should be as small as possible during the transformer design process, and in this way,Ls≫Lr, so duty ratio loss of three-level DC/DC controlled-source circuit after addition of commutation inductor is very small.


### 4.4. Selection of Flying Capacitor

When load current changes, the stability of a flying capacitor must be ensured. Primary current discharges the parallel capacitor through the flying capacitor during the turning off interval ofS1 and discharges parallel capacitor Cs1 during the turning off interval of S4. Under the steady state of the circuit, the flying capacitor is charged to Ui/2, and Ui/2 can be kept constant only when flying capacitance value is large enough, and capacitance value is related to the current flowing through it and expected voltage ripple, so capacitance value is as follows:(24)Css≥Ip0tdΔUCss,where td is the dead time and ΔUCss is the expected ripple voltage. In order to solve initial charging balance problem of the flying capacitor, a 470 k/2 W resistor is connected in parallel to it, and meanwhile, a 470 k/2 W resistor is in parallel connection with switching devices S1 and S4, which can compensate the initial imbalance of capacitance voltage.


## 4.1. Functions of Flying Capacitors and Flywheel Diodes

The operation process of full-bridge three-level DC/DC controlled-source circuit mentioned above has made a detailed analysis of flying capacitors and flywheel diodes at left-side bridge arm. Now functions of flying capacitors and flywheel diodes at right bridge arm will analyze. It can be known from driving waveforms in Figure7 that switching devices S5 and S6 (or S7 and S8) of right three-level bridge arm are simultaneously turned on and turned off; however, in the actual circuit, differences will certainly exist in turning on and turning off of S5 and S6 (or S7,S8). If there are no flying capacitors or flywheel diodes, it will possibly cause nonuniform voltage stress of switching devices. When switching device is turned on at ZVS, its antiparallel diodes have been turned on, and at the time, differences existing in turning on of switching devices will not cause nonuniformity of voltage stress of switching devices. Functions of flying capacitors and flywheel diodes will be mainly discussed under differences of turning off of switching devices.It is assumed that switching devicesS5 and S6 are simultaneously turned on and turned off; switching devices S7 and S8 will be simultaneously turned on but they cannot be simultaneously turned off. The discussion will be made under two circumstances: S7 is turned off before and after S8.


### 4.1.1. Switching DeviceS7 Is Turned Off before S8

Figure16 gives main waveforms when switching device S7 is turned off before S8. Before t0, switching devices S1, S2, S7, and S8 are turned on. At t0, switching devices S7 and S8 should be simultaneously turned off at first; the switching device S7 is turned off while S8 is still under on state due to the difference of characteristics of circuit devices. ip charges capacitor Cs7 and discharges the capacitor Cs6 through a flying capacitor Css2 and ip reduces.Figure 16
Main waveforms whenS7 is turned off before S8.Att1, S8 is turned off, and ip charges the capacitor Cs8 and discharges the capacitor Cs5 through the flying capacitor Css2. Meanwhile, ip continues to charge the capacitor Cs7 and discharge Cs6 through the flying capacitor Css2.(11)ip=iCs6+iCs7,(12)iCs5=iCs6+iCss2,(13)iCs8=iCs7−iCss2.Through the addition of formula (12) and (13), the following can be obtained by combining (11):(14)ip=iCs5+iCs8=iCs6+iCs7.It can be known from formula (14) that the current which discharges shunt capacitor of the lagging leg is ip and energy which realizes ZVS of the lagging leg is provided by commutation inductor.Att2, the voltage of the capacitor Cs6 reduces to zero if the primary current does not reduce to zero yet; the diode D6 is naturally turned on, and the switching device S6 can be turned on at zero voltage. ip continues to charge the capacitor Cs8 through a flying capacitor Css2 and discharge capacitor Cs5. At t3, the voltage of the capacitor Cs5 reduces to zero, if the primary current does not reduce to zero yet, the diode D5 will be naturally turned on, and switching device S5 can be turned on at zero voltage. As driving signals of S5 and S6 are the same, turning-on signals are sent to S5 and S6 at t3, which can be turned on at zero voltage.


### 4.1.2. Switching DeviceS7 Is Turned Off after S8

Figure17 gives main waveforms when switching device S7 is turned off after S8. Before t0, the switching device S8 is turned off while S7 is still turned on, primary current ip charges capacitor Cs7 and discharges the capacitor Cs5 through a flying capacitor Css2, and ip reduces. At t1, the switching device S7 is turned off, ip charges capacitor Cs7 and discharges the capacitor Cs6 through the flying capacitor Css2. Meanwhile, ip continues to charge Cs8 and discharge the capacitor Cs6 through the flying capacitor Css2. Formula (14) indicates that the current which discharges shunt capacitance of lagging leg is ip, and the energy which realizes ZVS of the lagging leg is provided by commutation inductor. At t2, the voltage of the capacitor Cs5 reduces to zero if the primary current does not decline to zero yet, the diode D5 is naturally turned on, and the switching device S5 can be turned on at zero voltage. ip continues to charge the capacitor Cs7 through a flying capacitor Css2 and discharge capacitor Cs7. At t3, the voltage of the capacitor Cs6 reduces to zero, the diode D6 is naturally turned on, and the switching device S6 can be turned on at zero voltage.Figure 17
Main waveforms whenS7 is turned off after S8.For right three-level bridge arm, if there is no flying capacitor or flywheel diode, voltage stress of the switching device which is turned off firstly will exceedUi/2 when the above two switching devices or below two switching devices are not simultaneously turned off, which will cause an imbalance of voltage stress borne by switching devices.The object of flywheel diode is to provide channel supplementing energy for flying capacitor and to make the voltage of the flying capacitor maintain atUi/2. It can be seen from the above analysis that flying capacitor not only realizes decoupling of charging and discharging processes of parallel capacitors of switching devices but also keeps voltage stress of switching devices at Ui/2, which avoids an imbalance of voltage stress caused by differences of turning on and turning off of two pairs of switching devices.


## 4.1.1. Switching DeviceS7 Is Turned Off before S8

Figure16 gives main waveforms when switching device S7 is turned off before S8. Before t0, switching devices S1, S2, S7, and S8 are turned on. At t0, switching devices S7 and S8 should be simultaneously turned off at first; the switching device S7 is turned off while S8 is still under on state due to the difference of characteristics of circuit devices. ip charges capacitor Cs7 and discharges the capacitor Cs6 through a flying capacitor Css2 and ip reduces.Figure 16
Main waveforms whenS7 is turned off before S8.Att1, S8 is turned off, and ip charges the capacitor Cs8 and discharges the capacitor Cs5 through the flying capacitor Css2. Meanwhile, ip continues to charge the capacitor Cs7 and discharge Cs6 through the flying capacitor Css2.(11)ip=iCs6+iCs7,(12)iCs5=iCs6+iCss2,(13)iCs8=iCs7−iCss2.Through the addition of formula (12) and (13), the following can be obtained by combining (11):(14)ip=iCs5+iCs8=iCs6+iCs7.It can be known from formula (14) that the current which discharges shunt capacitor of the lagging leg is ip and energy which realizes ZVS of the lagging leg is provided by commutation inductor.Att2, the voltage of the capacitor Cs6 reduces to zero if the primary current does not reduce to zero yet; the diode D6 is naturally turned on, and the switching device S6 can be turned on at zero voltage. ip continues to charge the capacitor Cs8 through a flying capacitor Css2 and discharge capacitor Cs5. At t3, the voltage of the capacitor Cs5 reduces to zero, if the primary current does not reduce to zero yet, the diode D5 will be naturally turned on, and switching device S5 can be turned on at zero voltage. As driving signals of S5 and S6 are the same, turning-on signals are sent to S5 and S6 at t3, which can be turned on at zero voltage.


## 4.1.2. Switching DeviceS7 Is Turned Off after S8

Figure17 gives main waveforms when switching device S7 is turned off after S8. Before t0, the switching device S8 is turned off while S7 is still turned on, primary current ip charges capacitor Cs7 and discharges the capacitor Cs5 through a flying capacitor Css2, and ip reduces. At t1, the switching device S7 is turned off, ip charges capacitor Cs7 and discharges the capacitor Cs6 through the flying capacitor Css2. Meanwhile, ip continues to charge Cs8 and discharge the capacitor Cs6 through the flying capacitor Css2. Formula (14) indicates that the current which discharges shunt capacitance of lagging leg is ip, and the energy which realizes ZVS of the lagging leg is provided by commutation inductor. At t2, the voltage of the capacitor Cs5 reduces to zero if the primary current does not decline to zero yet, the diode D5 is naturally turned on, and the switching device S5 can be turned on at zero voltage. ip continues to charge the capacitor Cs7 through a flying capacitor Css2 and discharge capacitor Cs7. At t3, the voltage of the capacitor Cs6 reduces to zero, the diode D6 is naturally turned on, and the switching device S6 can be turned on at zero voltage.Figure 17
Main waveforms whenS7 is turned off after S8.For right three-level bridge arm, if there is no flying capacitor or flywheel diode, voltage stress of the switching device which is turned off firstly will exceedUi/2 when the above two switching devices or below two switching devices are not simultaneously turned off, which will cause an imbalance of voltage stress borne by switching devices.The object of flywheel diode is to provide channel supplementing energy for flying capacitor and to make the voltage of the flying capacitor maintain atUi/2. It can be seen from the above analysis that flying capacitor not only realizes decoupling of charging and discharging processes of parallel capacitors of switching devices but also keeps voltage stress of switching devices at Ui/2, which avoids an imbalance of voltage stress caused by differences of turning on and turning off of two pairs of switching devices.


## 4.2. Conditions of Soft Switching

### 4.2.1. Soft Switching of the Leading Leg

It can be known from the previous analysis that to realize ZVS, the leading leg must have enough energy to take away electric charge on a parallel capacitorCs4 of switching device S4 and discharge parallel capacitor Cs1 of switching device S1 which is turned off; then, the energy used by the leading leg to realize ZVS is(15)Echop≥12Cs1Ui22+12Cs4Ui22=14CchopUi2.This energy mainly comes from primary converted value from load current. As output filter inductance is large enough, it is approximately believed that the current converted from constant current source to the primary side will easily let leading leg be turned on at zero voltage.


### 4.2.2. Soft Switching of Lagging Leg

Three-level DC/DC controlled-source circuit has three lagging devices which are simultaneously turned off. During the turning off process, ZVS of the lagging leg can be realized only by charging parallel capacitors on lagging device which is turned off and by completely discharging parallel capacitors on three lagging devices. Taking turning off ofS2, S7, and S8 as an example, they are divided into two stages as shown in Figure 3. In stage I, uAB reduces to zero, and corresponding to mode 3, energy is mainly provided by the current converted from output filter inductor in this stage:(16)Elag1≥12Cs2+Cs7+Cs8Udc32+12Cs3+Cs5+Cs6Udc62=524ClagUdc2.In Stage II,uAB rises to Ui, and corresponding to mode 4, energy is mainly provided by the current converted from commutation inductor in this stage.(17)Elag2≥12Cs2+Cs7+Cs8Udc22−Udc62+12Cs3+Cs5+Cs6Udc32=12ClagUdc2.The energy of the commutation inductor must satisfy the following expression:(18)12LcUdcTs4mLc2=12ClagUdc2.Then(19)Lc=Ts216m2Clag.If ZVS is realized when the output current is 1/5 of the designed full load, the duty cycle loss of conventional three-level converter is as follows:(20)Lc′=25Clagn2Udc2Io2.


## 4.2.1. Soft Switching of the Leading Leg

It can be known from the previous analysis that to realize ZVS, the leading leg must have enough energy to take away electric charge on a parallel capacitorCs4 of switching device S4 and discharge parallel capacitor Cs1 of switching device S1 which is turned off; then, the energy used by the leading leg to realize ZVS is(15)Echop≥12Cs1Ui22+12Cs4Ui22=14CchopUi2.This energy mainly comes from primary converted value from load current. As output filter inductance is large enough, it is approximately believed that the current converted from constant current source to the primary side will easily let leading leg be turned on at zero voltage.


## 4.2.2. Soft Switching of Lagging Leg

Three-level DC/DC controlled-source circuit has three lagging devices which are simultaneously turned off. During the turning off process, ZVS of the lagging leg can be realized only by charging parallel capacitors on lagging device which is turned off and by completely discharging parallel capacitors on three lagging devices. Taking turning off ofS2, S7, and S8 as an example, they are divided into two stages as shown in Figure 3. In stage I, uAB reduces to zero, and corresponding to mode 3, energy is mainly provided by the current converted from output filter inductor in this stage:(16)Elag1≥12Cs2+Cs7+Cs8Udc32+12Cs3+Cs5+Cs6Udc62=524ClagUdc2.In Stage II,uAB rises to Ui, and corresponding to mode 4, energy is mainly provided by the current converted from commutation inductor in this stage.(17)Elag2≥12Cs2+Cs7+Cs8Udc22−Udc62+12Cs3+Cs5+Cs6Udc32=12ClagUdc2.The energy of the commutation inductor must satisfy the following expression:(18)12LcUdcTs4mLc2=12ClagUdc2.Then(19)Lc=Ts216m2Clag.If ZVS is realized when the output current is 1/5 of the designed full load, the duty cycle loss of conventional three-level converter is as follows:(20)Lc′=25Clagn2Udc2Io2.


## 4.3. Duty Ratio Loss

Due to the existence of leakage inductance of the transformer, it will certainly bring about duty ratio loss. It can be known from previous analysis that inverted output voltageuAB has reached −Ui at t4, and this time is corresponding controlled duty ratio, but voltage us of secondary side of transformer does not rise to nUdc until t6, and this time is corresponding to effective duty ratio, and difference value between the two times is time of duty ratio loss as shown in Figure 3.(21)t45=Ipt4LrUi,t56=Ip0LrUi.Duty ratio loss is as follows:(22)Dloss=t45+t56Ts=2Ipt4+Ip0LrTsUi.The duty cycle loss of conventional three-level converter is as follows:(23)Dloss′=2Ipt4+Ip0LsTsUi.As commutation inductor is used to provide energy for turning off of lagging leg, leakage inductance should be as small as possible during the transformer design process, and in this way,Ls≫Lr, so duty ratio loss of three-level DC/DC controlled-source circuit after addition of commutation inductor is very small.


## 4.4. Selection of Flying Capacitor

When load current changes, the stability of a flying capacitor must be ensured. Primary current discharges the parallel capacitor through the flying capacitor during the turning off interval ofS1 and discharges parallel capacitor Cs1 during the turning off interval of S4. Under the steady state of the circuit, the flying capacitor is charged to Ui/2, and Ui/2 can be kept constant only when flying capacitance value is large enough, and capacitance value is related to the current flowing through it and expected voltage ripple, so capacitance value is as follows:(24)Css≥Ip0tdΔUCss,where td is the dead time and ΔUCss is the expected ripple voltage. In order to solve initial charging balance problem of the flying capacitor, a 470 k/2 W resistor is connected in parallel to it, and meanwhile, a 470 k/2 W resistor is in parallel connection with switching devices S1 and S4, which can compensate the initial imbalance of capacitance voltage.


## 5. Experimental Results

A prototype of a marine electromagnetic detection transmitter is used to verify the effectiveness of the contents stated above. Parameters are as follows: input voltageUi=540V, output voltage Uo=30V, output current Io=200A, and switching frequency fs=20kHz; commutation inductor: Lc=20μH; flying capacitor: Css=20μF; IGBT shunt capacitor: Cs=10nF; output filter inductor: L=40μH; output filter capacitor: C=1000μF, switching device: FF150R12RT4; and high-frequency rectifier diode: DPF240 × 200 NA. Figure 18 gives driving signals of four switching devices S1, S2, S3, S4 of the full-bridge three-level controlled-source circuit. S1 and S2 (or S4 and S3) are simultaneously turned on but with different turning off time. Output voltage and current are controlled by controlling pulse widths of S1, S4.Figure 18
Driving waveforms of full-bridge three-level controlled-source circuit.Figure19 gives waveforms of secondary rectifying voltage ud and commutation inductor current iLc of the high-frequency transformer. Commutation inductor current has different slopes under different output voltage values, but peak value point appears at converting the time of lagging leg, which solves difficult commutation problems of lagging leg and realizes soft switching of the controlled-source circuit within the whole power range.Figure 19
Waveforms of primary and secondary voltages of high-frequency transformer and commutation inductor current.Figures20 and 21 give waveforms of primary uAB and secondary rectifying voltage ud as well as primary currents ip of the high-frequency transformer in three-level DC/DC controlled-source circuit, respectively, under heavy load and light load. Under light load, switching devices also realize ZVS, which is identical to the theoretical analysis. Compared with two-level DC/DC controlled-source circuit, a level is added to secondary voltage, which results in a volume reduction of the output filter.Figure 20
Waveforms of primary and secondary voltages and current of transformer under heavy load.Figure 21
Waveforms of primary and secondary voltages and current of transformer under light load.Figure22 gives voltage waveforms at two ends of S1 and S2. Voltage borne by each switching device is half of the bus voltage, which greatly reduces voltage stress of switching device and solves large voltage stress problem of switching devices under high input voltage condition.Figure 22
Voltage waveforms at two ends ofS1 and S2.Figure23 shows the output voltage and current waveforms of the three-level controlled-source circuit under step response. It can be seen from the figure that the output voltage ripple is small, and the dynamic response is fast.Figure 23
Circuit testing efficiency.Experimental prototype of marine electromagnetic transmitter is shown in Figure24. The whole circuit is installed in a 23 cm (D) ∗ 780 cm (L) closed cylinder, with two ends connected with DC power supply and emitting electrode. Parameters include Vdc=540V, Vo=30V, Io=200A, and fs=20kHz. Lc=12μF, Lc′=20μF, m=20, n=16. Resonant inductor and commutation inductor are shown in Figure 25. Figure 26 gives change curves of three-level DC/DC controlled-source circuit efficiency with load change when resonant inductor and commutation reactor are used. Transmitter efficiency is improved, and maximum efficiency reaches 90% as the load increases. The main difference between the two lies in that when the load current is small, the transmitter using commutation inductor has higher efficiency.Figure 24
The physical picture of the marine transmitter.Figure 25
Resonant inductor and commutation inductor.
(a)
(b)Figure 26
Circuit testing efficiency.


## 6. Conclusions

For large voltage stress problem of switching devices in controlled-source circuit of marine electromagnetic detection transmitter under high input voltage condition, the three-level technology was used and voltage stress of switching device in the main circuit was reduced to half of the bus voltage, so this could reduce the rated voltage of switching device, improve circuit operation efficiency, and reduce transformer and filter volumes.In order to solve difficult current commutation problems of lagging legs, commutation inductor was added at the secondary side of the high-frequency transformer so as to realize soft switching of the controlled-source circuit within the whole power range and improve circuit efficiency. Moreover, this reduced requirement for leakage inductance of the high-frequency transformer and relieved transformer loss.On the condition that optimal control mode was satisfied, asymmetric phase-shift PWM control was introduced so that controlled duty ratio was corresponding to driving pulse width with definite physical significance, simple control, and convenience for the realization of digitalization.



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*Source: 3605604-2019-06-13.xml*