# Circularly Polarized Omnidirectional Antenna with Dipole Core and Diagonally Adjoined Parasitic Braces for ISM Band Applications

Authors: Piyapong Dangkham; Sitthichai Dentri; Chuwong Phongcharoenpanich; Prayoot Akkaraekthalin
Journal: International Journal of Antennas and Propagation
 (2019)

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## Abstract

This research proposes a circularly polarized (CP) single-fed omnidirectional dipole antenna operable in 2.45 GHz frequency for the industrial, scientific, and medical (ISM) radio band applications. The proposed antenna consisted of bisectional dipole core, a pair of quarter-wave baluns, and four diagonally adjoined parasitic braces. The bisectional dipole core was utilized to improve the antenna gain and realize omnidirectional radiation pattern, and the quarter-wave baluns were to symmetrize the current on the bisectional core. The four parasitic braces collectively generated circular polarization. In the study, simulations were conducted using CST Microwave Studio and a prototype antenna fabricated. To validate, experiments were carried out, and simulation and experimental results compared. The finding revealed good agreement between the simulation and experimental results. Essentially, in addition to achieving an antenna gain of 2.07 dBic, the proposed CP single-fed omnidirectional antenna is suited to ISM frequency band applications.


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## Body

## 1. Introduction

In modern wireless communication, transmitting and receiving antennas with omnidirectional radiation pattern are preferable. Nevertheless, omnidirectionality gives rise to multipath wave reflections and phase error in the receiving antenna. As a result, circular polarization is adopted to rectify the multipath effect.In theory, circular polarization of a patch antenna is realized by exciting two orthogonal components of identical amplitude [1]. Specifically, in [2], a circular monopolar patch antenna with U-slot patch and four slots achieved dual-band (2.45 and 5.8 GHz) CP omnidirectional radiation pattern at the respective gains of 1.37 and 4.37 dBi. In [3], a CP quad-band antenna using metamaterial was proposed. In [4], an antenna with four bended monopoles excited by feeding network was used to realize circular polarization with impedance bandwidth of 3.56% at 2.44 GHz and an average gain of 1.39 dBi. In [5], bended dipoles integrated with baluns were used to realize circular polarization with 2.32-2.61 GHz axial ratio bandwidth and a gain of 1.2 dBic. In [6], a circular slot fed by an L-shaped strip was proposed to realize circular polarization. The antenna gains in [2–6] were less than satisfactory, however.In [7], a quarter-wave shorted patch connected with two-PCB strip was utilized to generate circular polarization with impedance bandwidth of 4.3% at 5.8 GHz. In [8], a weak-coupling mechanism was incorporated into feeding network to realize high gain and wide 3-dB AR bandwidth. In [9], a 2.4 GHz two-faced slot CP antenna was proposed to realize circular polarization. In [10], two off-center-fed dipoles were proposed to realize a broadband CP antenna.In [11], a CP antenna with rotatable dipole-shaped radiation pattern achieved impedance bandwidth of 2.4-2.51 GHz with a gain of 1.9 dBic. In [12], circular polarization was realized using a circular patch with six vortex slots and six shorting pins. In [13], four arcs were incorporated to improve circular polarization of a patch antenna. In [14], elliptical-ring slot was deployed surrounding the main patch to improve circular polarization. In [15], a circular patch connected to ground plane by conductive vias achieved wideband CP but low antenna gain. The antennas in [11–15] required multilayered PCB and shorting vias.In [16], a dielectric resonator antenna (DRA) achieved CP omnidirectional radiation pattern. In [17], a rectangular DRA above the ground plane could achieve dual frequency of 1.58 GHz and 2.4 GHz. In [18], four 30°-rotated rectangular dielectric layers generated 90°-phase current difference between layers, thus achieving circular polarization. In [19], a DRA could achieve CP radiation pattern with impedance matching of -40 dB. In [20], a 2.4 GHz CP antenna with top-loaded Alford loop was proposed. In [21], a CP DRA fed by microstrip line achieved a 3-dB AR bandwidth of 85 MHz at 2.45 GHz frequency. However, the dielectric resonator antennas in [16–21] suffered from fabrication complexity.In [22–24], omnidirectional dipole antennas could achieve improved impedance bandwidth and high antenna gains. However, they failed to realize circular polarization. In [25], a CP array antenna with parallel striplines achieved 3-dB AR between 890 and 930 MHz with omnidirectional radiation pattern. In [26], a feeding probe and parasitic dielectric paralleled pipe element achieved a 3-dB AR bandwidth of 54.9% with omnidirectional radiation pattern. In [27], a torus-knot antenna could achieve CP omnidirectional beam. In [28], an antenna using two orthogonally aligned circles achieved a 3-dB AR bandwidth of 58% with omnidirectional radiation pattern.Specifically, the aim of this research is to propose a circularly polarized single-fed omnidirectional dipole antenna operable in 2.45 GHz center frequency for ISM band applications. The proposed CP omnidirectional antenna was comprised of bisectional dipole core, a pair of quarter-wave baluns, and four diagonally adjoined parasitic braces. The bisectional core (upper and lower core sections) was deployed to enhance the antenna gain and realize omnidirectional radiation pattern, while the baluns symmetrized the current on the upper and lower core sections. Circular polarization was realized by four diagonally adjoined parasitic braces. Simulations were performed using CST Microwave Studio [29] and an antenna prototype fabricated. Experiments were subsequently carried out and simulation and experimental results compared, including matching impedance, the 3-dB axial ratio bandwidth, gain, and 3-dB axial ratio beamwidth.


## 2. Antenna Structure

Figures1(a) and 1(b) illustrate the geometry of the circularly polarized (CP) single-fed omnidirectional dipole antenna, consisting of bisectional dipole core, a pair of quarter-wave baluns, and four diagonally adjoined parasitic braces. The CP omnidirectional antenna was 52×52×56 mm in size. The copper dipole core was of two sections: upper and lower sections, whose distance (
δ
) was 2 mm. The radius (
r
d
) and thickness of the dipole core were 10 and 1 mm. The length of each core section (
L
) was 27 mm and perpendicular to the quarter-wave baluns. The bisectional core was connected to a coaxial feed via balun. The bisectional dipole core was utilized to obtain omnidirectional radiation pattern. Meanwhile, the parasitic braces were incorporated to generate circular polarization.Figure 1
The circularly polarized single-fed omnidirectional dipole antenna: (a) geometry, (b) configuration with balun, (c) prototype.
(a)
(b)
(c)Each parasitic brace resembled two diagonally adjoined rectangular copper plates of identical size, with 17.5, 16, and 1 mm in height(
H
), width (
W
), and thickness. The adjoining region between both rectangular plates (
v
2
) was 1 mm2. There were four diagonally adjoined parasitic braces enclosing the bisectional core. The void space between the parasitic braces and core center was 26 mm in distance orλ/5 whereλ is the wavelength at the center frequency (2.45 GHz). The parasitic braces were individually placed at an angle (
α
) of 90° in relation to adjacent braces to generate circular polarization. Interestingly, the thickness of parasitic braces beyond 1 mm shifted the resonant frequency of impedance matching (
S
11
) and axial ratio (AR) to lower frequencies, thereby worsening S
11.Figure1(b) depicts the two copper quarter-wave baluns placed in parallel (2 mm apart) and individually connected to the bisectional core. The balun was 100, 5, and 1 mm in length, diameter, and thickness. The baluns were then shorted at a distance of 30 mm from the dipole core. The pair of baluns were utilized to induce current symmetry on the bisectional core.Figure1(c) shows a prototype of the 2.45 GHz CP single-fed omnidirectional dipole antenna. The bisectional dipole core individually connected to the two copper baluns. The 50-Ω SMA-type coaxial cable was used to feed the signal. The core and ground structure of the coaxial cable were individually connected to the bisectional dipole core through the balun. Two acrylic discs individually attached to the upper and lower sections of the dipole core. The parasitic braces adhered to both of the acrylic discs along their curves and made the bisectional core apart. Table 1 tabulates the optimal parameters of the proposed CP single-fed omnidirectional antenna operable at 2.45 GHz center frequency.Table 1
Parameters of the CP omnidirectional dipole antenna operable at 2.45 GHz.
Parameter
Description
Electrical size
Physical size
L
Length of sectional core (single section)
0.22λ
27.0 mm
δ
Distance between bi-sectional cores
0.02λ
2.0 mm
r
d
Radius of bi-sectional core
0.08λ
10.0 mm
W
Width of rectangular plate
0.13λ
16.0 mm
H
Height of rectangular plate
0.14λ
17.5 mm
v
Adjoining region of two rectangular plates
0.01λ
1.0 mm
r
p
Distance between parasitic braces and core center
0.21λ
26.0 mm
α
Angle between two adjacent plates
90.0°


## 3. Simulation and Measurement Results

The antenna prototype was then experimented and results were compared with simulation results. Figure2(a) compares the simulation and measured S
11. The simulation and measured S
11 (<-10 dB) covered 2.37-2.59 GHz (8.87%) and 2.35-2.57 GHz (8.94%), respectively, encompassing the ISM frequency band of 2.4-2.484 GHz. In Figure 2(b), the simulation and measured realized gains over the ISM band were 1.98 and 2.07 dBic. Meanwhile, the simulation and measured 3-dB AR bandwidth at the center frequency were 2.40-2.55 GHz (6.06%) and 2.40-2.60 GHz (8.00%).Figure 2
Simulation and measured results of the CP single-fed omnidirectional dipole antenna: (a)S
11, (b) AR and gain.
(a)
(b)Figures3(a) and 3(b), respectively, illustrate the simulation and measured AR in thex-z andy-z planes at 2.45 GHz. The simulation and measured AR of bothx-z andy-z planes were symmetrical and of right-hand circular polarization (RHCP). In Figure 3(a), the simulation and measured axial ratio beamwidth (AR-BW) in thex-zplane were 54° and 65°, respectively. The corresponding AR-beamwidths in they-z plane were identical (i.e., 360°).Figure 3
Simulation and measured AR of the CP single-fed omnidirectional dipole antenna at 2.45 GHz: (a)x-z, (b)y-z plane.
(a)
(b)Figures4(a) and 4(b), respectively, illustrate the simulation and measured radiation patterns in thex-z andy-z planes at 2.45 GHz. The radiation patterns were symmetrical and omnidirectional with RHCP. In thex-zplane, the simulation and measured AR-beamwidth were 74° and 75°, respectively, and the corresponding AR-beamwidths in they-z plane were also identical (360°).Figure 4
Simulation and measured radiation patterns of the CP single-fed omnidirectional dipole antenna at 2.45 GHz: (a)x-z, (b)y-z plane.
(a)
(b)Table2 compares the simulated and measured S
11, AR, half-power beamwidth, AR-BW, gain, and polarization at 2.45 GHz operating frequency. The simulation and experimental results exhibited good agreement.Table 2
Characteristics of CP single-fed omnidirectional antenna operable at 2.45 GHz.
Characteristics
Simulation
Measured
S
11
2.37 – 2.59 GHz
2.35 – 2.57 GHz
AR
2.40 – 2.55 GHz
2.40 – 2.60 GHz
HPBW
x–z
74°
75°
y–z
360°
360°
3-dB AR-BW
x–z
54°
65°
y–z
360°
360°
Gain
1.98 dBic
2.07 dBic
Polarization
RHCP
RHCP


## 4. Design, Parametric Study, and Analysis

### 4.1. Evolutionary Stages of Proposed Antenna

Figure5 illustrates the four evolutionary stages of the antenna, including conventional 6mm-Ø dipole core without parasitic brace (scheme A), with single parasitic brace (scheme B) and four parasitic braces (scheme C), and expanded 20mm-Ø dipole core with four parasitic braces (scheme D). Figures 6(a) and 6(b), respectively, illustrate the simulated S
11 and AR under scheme A, scheme B, scheme C, and scheme D using CST Microwave Studio.Figure 5
The evolutionary stages of CP single-fed omnidirectional antenna.Figure 6
The evolutionary stages of CP single-fed omnidirectional antenna: (a) simulatedS
11, (b) AR.
(a)
(b)In schemes A and B, the simulatedS
11 was below -10 dB (S
11
<
-
10
d
B), but both schemes failed to achieve circular polarization (AR = 40 dB and 27 dB for schemes A and B). In scheme C, the antenna failed to achieve impedance matching (S
11
>
-
10
d
B) despite AR approaching 3.0 dB. Meanwhile, scheme D achieved S
11 and circular polarization (AR<3.0 dB) at the 2.45 GHz center frequency. The optimal radius of the dipole core (r
d) with four parasitic braces was thus 10 mm.


### 4.2. Length of One Single Section of the Dipole Core(
L
)

Figures7(a)–7(c), respectively, illustrate the simulated S
11, AR, gain, and half power beamwidth (HPBW) under variable lengths of dipole core (single section): 21, 24, 27, 30, and 33 mm. The findings revealed that, with increase in L
, the resonant frequency of S
11 became lower while that of AR became minimally higher. The effects were expected because it is well known that resonant frequency decreases with an increase in the length of dipole core. The optimal L was 27 mm, where S
11 < -10 dB and AR < 3 dB at 2.45 GHz frequency. In Figure 7(c), the maximum gain was in the vicinity of the 2dBic standard dipole, givenL between 27 and 30 mm, which is the vicinity of half-wave dipole core. Meanwhile, the HPBW ofx-z plane decreased with increase in the length of dipole core. The HPBW of the optimalL (27 mm) was thus 74 degrees.Figure 7
Simulation results under variable lengths (L) of dipole core (single section): (a) S
11, (b) AR, (c) gain and HPBW.
(a)
(b)
(c)


### 4.3. Distance between Upper and Lower Core Sections(
δ
)

Figures8(a)–8(c), respectively, illustrate the simulated S
11, AR, gain, and HPBW under variable distances between two sections of the dipole core (
δ
): 1.0, 1.5, 2.0, 2.5, and 3.0 mm. The findings showed that δ below or above 2.0 mm resulted in the resonant frequency of S
11 falling outside the target operating frequency (2.45 GHz), while variation in δ had negligible impact on AR. Since δ had no effect on AR, it could be used to tune the resonant frequency of S
11 without the worry about the 3-dB AR. The optimal δ was thus 2 mm. In Figure 8(c), with δ below or above 2.0 mm, the gain became lower while variation in δ had negligible impact on HPBW because the length of dipole core was not changed.Figure 8
Simulation results under variable distances between two core sections(
δ
): (a) S
11, (b) AR, (c) gain and HPBW.
(a)
(b)
(c)


### 4.4. Radius of Bisectional Dipole Core(
r
d
)

Figures9(a)–9(c), respectively, depict the simulated S
11, AR, gain, and HPBW under variable radii of bisectional dipole core (r
d): 3.0, 6.5, 10.0, 13.5, and 17.0 mm. In Figure 9(a), r
d below or above 10.0 mm gave rise to the resonant frequency of S
11 falling outside the center frequency (2.45 GHz). Meanwhile, the resonant frequency of AR increased with increase in the core radius. The S
11 and AR resonant frequencies were sensitive to r
d. The optimal r
d was thus 10.0 mm. In Figure 9(c), r
d below or above 10.0 mm gave worse to the gain. Meanwhile, the HPBW increased with increase in the core radius.Figure 9
Simulation results under variable radii of the dipole core (r
d): (a) S
11, (b) AR, (c) gain and HPBW.
(a)
(b)
(c)


### 4.5. Width of Rectangular Parasitic Plate(
W
)

Figures10(a)–10(c), respectively, illustrate the simulated S
11, AR, gain, and HPBW under variable rectangular parasitic plate widths (
W
): 13.0, 14.5, 16.0, 17.5, and 19.0 mm. As previously discussed, four diagonally adjoined parasitic braces were deployed to realize circular polarization at the 2.45 GHz center frequency. The simulation results indicated that the resonant frequency of S
11 and AR approached the center frequency (2.45 GHz) as W increased. However, W beyond 16.0 mm adversely affected S
11 and AR resonant frequencies because they directly affected the field coupling between the bisectional dipole core and the parasitic plates. It was found that resonant frequency decreases with an increase in the width of rectangular parasitic plate. The optimal W was thus 16.0 mm. In Figure 10(c), W below or above 16.0 mm gave worse to the gain. The HPBW increased with increase in the width of rectangular parasitic plate between 14.5 and 17.5 mm.Figure 10
Simulation results under variable rectangular plate widths(
W
): (a) S
11, (b) AR, (c) gain and HPBW.
(a)
(b)
(c)


### 4.6. Length of Rectangular Parasitic Plate(
H
)

Figures11(a)–11(c), respectively, show the simulated S
11, AR, gain, and HPBW under variable rectangular parasitic plate lengths (
H
): 11.5, 14.5, 17.5, 20.5, and 23.5 mm. The findings revealed that the resonant frequency of S
11 and AR approached the target frequency (2.45 GHz) as H increased. Nonetheless, H beyond 17.5 mm shifted S
11 and AR resonant frequencies below the target frequency. It was found that the effects were similar to those of W. The optimal H was thus 17.5 mm. In Figure 11(c), H below or above 17.5 mm gave worse to the gain. The HPBW increased with increase in the length of rectangular parasitic plate between 17.5 and 23.5 mm, while variation inHbetween 11.5 and 17.5 mm had negligible impact on HPBW.Figure 11
Simulation results under variable rectangular plate lengths (H): (a) S
11, (b) AR, (c) gain and HPBW.
(a)
(b)
(c)


### 4.7. Distance between Parasitic Brace and the Core Center(
r
p
)

Figures12(a)–12(c), respectively, illustrate the simulated S
11, AR, gain, and HPBW under variable distances between parasitic brace and the core center (r
p): 22, 24, 26, 28, and 30 mm. The simulation results showed that the resonant frequency of S
11 approached the target center frequency (2.45 GHz) as r
p increased. However, beyond the 26mm threshold, the S
11 resonant frequency was below the target frequency. The AR resonant frequency was in the vicinity of the target frequency, given r
p between 24 and 30 mm. The resonant frequency of AR was below the target frequency for r
p of 22 mm. The adjustment of r
p was less affected than the adjustment of r
d. The optimal r
p was 26 mm. The variation in r
p had negligible impact on the gain. Meanwhile, the HPBW decreased with increase in the distance between parasitic brace and the core center.Figure 12
Simulation results under variable distances between parasitic brace and the core center (r
p): (a) S
11, (b) AR, (c) gain and HPBW.
(a)
(b)
(c)


### 4.8. Current Vector Distribution on Diagonally Adjoined Parasitic Brace

Figure13 illustrates the simulated current vector distribution on a single diagonally adjoined parasitic brace relative to time (
t
). The magnitude of current vector was identical with 90° phase difference, independent of time. At 2.45 GHz, the direction of surface current was counterclockwise as time (
t
) shifted from 0, T/2, T/4, to 3T/4. The collective use of four diagonally adjoined parasitic braces thus generated circular polarization.Figure 13
Current vector distribution on a single diagonally adjoined parasitic brace.


## 4.1. Evolutionary Stages of Proposed Antenna

Figure5 illustrates the four evolutionary stages of the antenna, including conventional 6mm-Ø dipole core without parasitic brace (scheme A), with single parasitic brace (scheme B) and four parasitic braces (scheme C), and expanded 20mm-Ø dipole core with four parasitic braces (scheme D). Figures 6(a) and 6(b), respectively, illustrate the simulated S
11 and AR under scheme A, scheme B, scheme C, and scheme D using CST Microwave Studio.Figure 5
The evolutionary stages of CP single-fed omnidirectional antenna.Figure 6
The evolutionary stages of CP single-fed omnidirectional antenna: (a) simulatedS
11, (b) AR.
(a)
(b)In schemes A and B, the simulatedS
11 was below -10 dB (S
11
<
-
10
d
B), but both schemes failed to achieve circular polarization (AR = 40 dB and 27 dB for schemes A and B). In scheme C, the antenna failed to achieve impedance matching (S
11
>
-
10
d
B) despite AR approaching 3.0 dB. Meanwhile, scheme D achieved S
11 and circular polarization (AR<3.0 dB) at the 2.45 GHz center frequency. The optimal radius of the dipole core (r
d) with four parasitic braces was thus 10 mm.


## 4.2. Length of One Single Section of the Dipole Core(
L
)

Figures7(a)–7(c), respectively, illustrate the simulated S
11, AR, gain, and half power beamwidth (HPBW) under variable lengths of dipole core (single section): 21, 24, 27, 30, and 33 mm. The findings revealed that, with increase in L
, the resonant frequency of S
11 became lower while that of AR became minimally higher. The effects were expected because it is well known that resonant frequency decreases with an increase in the length of dipole core. The optimal L was 27 mm, where S
11 < -10 dB and AR < 3 dB at 2.45 GHz frequency. In Figure 7(c), the maximum gain was in the vicinity of the 2dBic standard dipole, givenL between 27 and 30 mm, which is the vicinity of half-wave dipole core. Meanwhile, the HPBW ofx-z plane decreased with increase in the length of dipole core. The HPBW of the optimalL (27 mm) was thus 74 degrees.Figure 7
Simulation results under variable lengths (L) of dipole core (single section): (a) S
11, (b) AR, (c) gain and HPBW.
(a)
(b)
(c)


## 4.3. Distance between Upper and Lower Core Sections(
δ
)

Figures8(a)–8(c), respectively, illustrate the simulated S
11, AR, gain, and HPBW under variable distances between two sections of the dipole core (
δ
): 1.0, 1.5, 2.0, 2.5, and 3.0 mm. The findings showed that δ below or above 2.0 mm resulted in the resonant frequency of S
11 falling outside the target operating frequency (2.45 GHz), while variation in δ had negligible impact on AR. Since δ had no effect on AR, it could be used to tune the resonant frequency of S
11 without the worry about the 3-dB AR. The optimal δ was thus 2 mm. In Figure 8(c), with δ below or above 2.0 mm, the gain became lower while variation in δ had negligible impact on HPBW because the length of dipole core was not changed.Figure 8
Simulation results under variable distances between two core sections(
δ
): (a) S
11, (b) AR, (c) gain and HPBW.
(a)
(b)
(c)


## 4.4. Radius of Bisectional Dipole Core(
r
d
)

Figures9(a)–9(c), respectively, depict the simulated S
11, AR, gain, and HPBW under variable radii of bisectional dipole core (r
d): 3.0, 6.5, 10.0, 13.5, and 17.0 mm. In Figure 9(a), r
d below or above 10.0 mm gave rise to the resonant frequency of S
11 falling outside the center frequency (2.45 GHz). Meanwhile, the resonant frequency of AR increased with increase in the core radius. The S
11 and AR resonant frequencies were sensitive to r
d. The optimal r
d was thus 10.0 mm. In Figure 9(c), r
d below or above 10.0 mm gave worse to the gain. Meanwhile, the HPBW increased with increase in the core radius.Figure 9
Simulation results under variable radii of the dipole core (r
d): (a) S
11, (b) AR, (c) gain and HPBW.
(a)
(b)
(c)


## 4.5. Width of Rectangular Parasitic Plate(
W
)

Figures10(a)–10(c), respectively, illustrate the simulated S
11, AR, gain, and HPBW under variable rectangular parasitic plate widths (
W
): 13.0, 14.5, 16.0, 17.5, and 19.0 mm. As previously discussed, four diagonally adjoined parasitic braces were deployed to realize circular polarization at the 2.45 GHz center frequency. The simulation results indicated that the resonant frequency of S
11 and AR approached the center frequency (2.45 GHz) as W increased. However, W beyond 16.0 mm adversely affected S
11 and AR resonant frequencies because they directly affected the field coupling between the bisectional dipole core and the parasitic plates. It was found that resonant frequency decreases with an increase in the width of rectangular parasitic plate. The optimal W was thus 16.0 mm. In Figure 10(c), W below or above 16.0 mm gave worse to the gain. The HPBW increased with increase in the width of rectangular parasitic plate between 14.5 and 17.5 mm.Figure 10
Simulation results under variable rectangular plate widths(
W
): (a) S
11, (b) AR, (c) gain and HPBW.
(a)
(b)
(c)


## 4.6. Length of Rectangular Parasitic Plate(
H
)

Figures11(a)–11(c), respectively, show the simulated S
11, AR, gain, and HPBW under variable rectangular parasitic plate lengths (
H
): 11.5, 14.5, 17.5, 20.5, and 23.5 mm. The findings revealed that the resonant frequency of S
11 and AR approached the target frequency (2.45 GHz) as H increased. Nonetheless, H beyond 17.5 mm shifted S
11 and AR resonant frequencies below the target frequency. It was found that the effects were similar to those of W. The optimal H was thus 17.5 mm. In Figure 11(c), H below or above 17.5 mm gave worse to the gain. The HPBW increased with increase in the length of rectangular parasitic plate between 17.5 and 23.5 mm, while variation inHbetween 11.5 and 17.5 mm had negligible impact on HPBW.Figure 11
Simulation results under variable rectangular plate lengths (H): (a) S
11, (b) AR, (c) gain and HPBW.
(a)
(b)
(c)


## 4.7. Distance between Parasitic Brace and the Core Center(
r
p
)

Figures12(a)–12(c), respectively, illustrate the simulated S
11, AR, gain, and HPBW under variable distances between parasitic brace and the core center (r
p): 22, 24, 26, 28, and 30 mm. The simulation results showed that the resonant frequency of S
11 approached the target center frequency (2.45 GHz) as r
p increased. However, beyond the 26mm threshold, the S
11 resonant frequency was below the target frequency. The AR resonant frequency was in the vicinity of the target frequency, given r
p between 24 and 30 mm. The resonant frequency of AR was below the target frequency for r
p of 22 mm. The adjustment of r
p was less affected than the adjustment of r
d. The optimal r
p was 26 mm. The variation in r
p had negligible impact on the gain. Meanwhile, the HPBW decreased with increase in the distance between parasitic brace and the core center.Figure 12
Simulation results under variable distances between parasitic brace and the core center (r
p): (a) S
11, (b) AR, (c) gain and HPBW.
(a)
(b)
(c)


## 4.8. Current Vector Distribution on Diagonally Adjoined Parasitic Brace

Figure13 illustrates the simulated current vector distribution on a single diagonally adjoined parasitic brace relative to time (
t
). The magnitude of current vector was identical with 90° phase difference, independent of time. At 2.45 GHz, the direction of surface current was counterclockwise as time (
t
) shifted from 0, T/2, T/4, to 3T/4. The collective use of four diagonally adjoined parasitic braces thus generated circular polarization.Figure 13
Current vector distribution on a single diagonally adjoined parasitic brace.


## 5. Comparison between CP Omnidirectional Antennas

The overall dimension of the proposed antenna is 52 mm × 52 mm × 56 mm, excluding the baluns and the coaxial connector. The baluns are necessary to symmetrize the current on the bisectional core and shorted at a distance of 30 mm from the bisectional core. The long baluns are used to support the measurement setup. For comparison of the various performance, some antennas in the references are not radiating CP with the omnidirectional radiation pattern, and some antennas operate outside 2.4-2.484 GHz. Thus, the related CP omnidirectional antennas encompassing the ISM frequency band of 2.4-2.484 GHz are listed in Table3. In the comparison, the antennas in [5, 16] are smaller than the proposed antenna, but it achieved a low gain. Moreover, the feeding network was required in [5]. The antenna in [11] achieved a similar gain to the proposed antenna, but it suffered from bulkiness. The antenna in [15] has the highest gain, but its size is the biggest (more than 3 times compared with the proposed antenna). Meanwhile, the antenna in [19, 20] achieved a similar size to the proposed antenna, but their gains were lower, and they used multiport feeding. For antennas operating in 2.4-2.484 GHz band, the proposed antenna possesses compromised performance because it has the high gain with the compact size. The advantages of the proposed antenna are that they possessed a similar gain to the standard dipole antenna, a single-fed antenna, and no feeding network was required in the antenna structure.Table 3
Comparison between CP omnidirectional antennas.
Ref.
Impedance bandwidth
AR bandwidth
Gain
Size (mm3)
Radiation Pattern
5
2.32 – 2.54 GHz
2.32 – 2.61 GHz
1.20 dBic
33 × 33 × 20
Omnidirectional
11
2.40 – 2.51 GHz
2.46 – 2.48 GHz
1.90 dBic
80 × 46 × 6
Omnidirectional
15
2.27 – 2.77 GHz
2.25 – 2.73 GHz
3.75 dBic
180 × 180 × 3
Omnidirectional
16
2.30 – 2.94 GHz
2.39 – 2.57 GHz
0.91 dBic
39 × 39 × 33
Omnidirectional
19
2.37 – 2.63 GHz
2.37 – 2.56 GHz
1.56 dBic
53 × 53 × 20
Omnidirectional
20
2.39 – 2.51 GHz
2.39 – 2.51 GHz
1.12 dBic
50 × 50 × 20
Omnidirectional
proposed
2.35 – 2.57 GHz
2.40 – 2.60 GHz
2.07 dBic
52 × 52 × 56
Omnidirectional


## 6. Conclusion

This research proposed a 2.45 GHz circularly polarized single-fed omnidirectional dipole antenna for ISM frequency band applications. The CP omnidirectional antenna consisted of bisectional dipole core (upper and lower sections), a pair of quarter-wave baluns, and four diagonally adjoined parasitic braces. The bisectional dipole core was utilized to enhance the antenna gain and realize omnidirectional radiation pattern, and the quarter-wave baluns were to symmetrize the current on the upper and lower core sections. The four parasitic braces were incorporated to induce circular polarization. The measuredS
11 bandwidth, 3-dB AR bandwidth, gain, and 3-dB AR beamwidth inx-z plane were 2.35-2.57 GHz (8.94%), 2.40-2.60 GHz (8.00%), 2.07 dBic, and 65°. In essence, the proposed CP omnidirectional antenna could achieve a high antenna gain and is suited to ISM frequency band applications.



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*Source: 2463871-2019-08-21.xml*