# A Dual-Band SiGe HBT Frequency-Tunable and Phase-Shifting Differential Amplifier Employing Varactor-Loaded, Stacked LC Resonators

Authors: Kazuyoshi Sakamoto; Yasushi Itoh
Journal: International Journal of Microwave Science and Technology
 (2012)

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## Abstract

A dual-band SiGe HBT frequency-tunable and phase-shifting differential amplifier has been developed for the future active phased array antennas with a multiband, multibeam, and multitarget tracking operation. The amplifier uses varactor-loaded, stacked LC resonators in the design of the output circuit in order to provide frequency-tunable and phase-shifting capabilities for dual frequencies. By utilizing the varactor-loaded LC resonator, which has a variable resonant frequency and a large insertion phase variation, frequency-tunable and phase-shifting performances become available. Moreover, by using the stacked configuration, the frequency and insertion phase can be varied independently for dual frequencies. A dual-band SiGe HBT differential amplifier has achieved a lower-frequency tuning range of 0.56 to 0.7 GHz for a higher fixed frequency of 0.97 GHz as well as a higher-frequency tuning range of 0.92 to 1.01 GHz for a lower fixed frequency of 0.63 GHz. A lower-frequency phase variation of 99° and a higher-frequency phase variation of 90.3° have been accomplished at 0.63 and 0.97 GHz, respectively. This is the first report on the dual-band differential amplifier with frequency-tunable and phase-shifting capabilities.


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## Body

## 1. Introduction

Recent and future wireless systems require a wide range of data rates over several frequency bands, and thus adaptive and reconfigurable transceivers become necessary to support high performance and flexibility [1]. Meanwhile, microwave and millimeter-wave sensors and radar systems require multiband, multibeam scanning phased array antennas as well as a multi-target tracking capability [2]. To address these requirements, multifunction capabilities are strongly required. Especially multiband amplification, frequency tunability, gain and phase control are crucial for realizing adaptive and reconfigurable phased array systems. Most of the traditional multiband amplifiers, however, provide a single function [3]. Meanwhile, the authors have presented multiband amplifiers with frequency-tunable as well as gain control capabilities for bandpass and bandstop types [4, 5]. To enhance the operational capability, a novel design approach is proposed in this paper for the multiband low-noise differential amplifier with frequency-tunable and phase-shifting capabilities. In order to realize both frequency-tunable and phase-shifting performances, varactor-loaded LC resonators are used in the design of the output circuit of the differential amplifier. LC resonators are widely used in the reflection type phase shifter because they provide a full 360° phase shift [6–8]. With the use of the parallel LC resonator in the output shunt circuit, frequency-tunable and phase-shifting performances with high gain can be achieved at around resonant frequencies. In addition, by cascading the parallel LC resonators in a stacked form, frequency and insertion phases can be varied independently for multiple frequencies. As a multiband phase shifter, the resonator-switching or diplexer method has been proposed for the loaded-line type [9], reflection type [10], and LPF/HPF type [11]. These phase shifters, however, produce additional losses due to the switch or diplexer. Since the stacked LC resonator does not require additional switches or diplexers, a low insertion loss and a large phase variation can be expected. To make sure the usefulness of the novel design approach proposed in this paper, the dual-band SiGe HBT differential amplifier with frequency-tunable and phase-shifting capabilities has been actually designed, fabricated, and tested.


## 2. Circuit Design

A schematic diagram of the dual-band frequency-tunable and phase-shifting differential amplifier is shown in Figure1. It employs varactor-loaded, stacked LC resonators in the design of the output circuit to achieve frequency-tunable and phase-shifting capabilities for dual frequencies. L1, L2, C1, and C2 are inductors and capacitors consisting of LC resonators. CV1 and CV2 are a variable capacitance of the varactor diode which is controlled by VC1 and VC2. CC is a decoupling capacitor having a large capacitance value. Resonant frequencies f1 and f2 are given as follows:
(1)f1=12πL1(C1+CV1),(2)f2=12πL2(C2+CV2).Figure 1
Schematic diagram of the dual-band frequency-tunable and phase-shifting differential amplifier.When two parallel LC resonators are cascaded in a stacked form, the following bandstop frequencyf3 is generated. Consider
(3)f3=12π1/L1+1/L2C1+CV1+C2+CV2.Nowf1<f2 (L1C1>L2C2) then f3 is necessarily sandwiched between f1 and f2 as
(4)f1<f3<f2.C
V
1 and CV2 are carefully designed in accordance with the target frequency range and phase-shifting amount. In order to realize CV1 and CV2, several varactor diodes with a fixed capacitance ratio are connected in series or parallel.A locus of the reflection coefficientΓL of the varactor-loaded, stacked LC resonator is graphically shown in Figure 2. As the frequency increases from zero to f1, ΓL moves from the short to the open-circuited point P. Since the LC resonator is actually lossy, the point P has finite impedance. Then as the frequency increases up to f3, ΓL moves to the series resonant point q where the impedance is not ideally short due to the lossy circuit. As the frequency increases from f3 to f2, ΓL comes back to the point P again. Finally ΓL goes to short. At around the resonant frequencies f1 and f2, the insertion phase drastically varies from +180° to −180°. Therefore, with the precise control of VC1 and VC2, the insertion phase can be tuned. In addition, at the resonant frequencies f1 and f2, a peak gain of the amplifier can be obtained. The mechanism is depicted graphically in Figure 3 by using a frequency response of the amplifier.Figure 2
Locus of the reflection coefficientΓL of the varactor-loaded, stacked LC resonator.Figure 3
Frequency response of the dual-band frequency-tunable and phase-shifting differential amplifier.It is clearly demonstrated that the frequency can be varied with the coarse control ofVC1 and VC2 and that the insertion phase can be also varied with the fine control of VC1 and VC2. f1, f2, and f3 are calculated by using from (1) to (3) for the circuit element values listed in Table 1. The circuit element values were chosen to meet the condition of (4). When CV1 varies from 6 to 15 pF, CV2 has a fixed value of 6 pF. Meanwhile, when CV2 varies from 6 to 15 pF, CV1 has a fixed value of 15 pF. The calculated results are shown in Figures 4 and 5, respectively. It is noted that f1 and f2 can be tuned independently and that f3 is necessarily sandwiched between f1 andf2.Table 1
Circuit element values of Figure1.
Element
Value
L
1 [nH]
1
C
1 [pF]
10
C
V
1 [pF]
6–15
L
2 [nH]
1
C
2 [pF]
3
C
V
2 [pF]
6–15
C
C [pF]
1000Figure 4
Calculatedf1, f2, and f3 (CV1 varies from 6 to 15 pF. CV2 has a fixed value of 6 pF).Figure 5
Calculatedf1, f2, and f3 (CV2 varies from 6 to 15 pF. CV1 has a fixed value of 15 pF).


## 3. Circuit Fabrication

A photograph of the dual-band frequency-tunable and phase-shifting differential amplifier is shown in Figure6.Figure 6
Photograph of the dual-band frequency-tunable and phase-shifting differential amplifier: 16 × 16 × 1.2 mm3.The differential amplifier was fabricated on the FR-4 substrate with a dielectric constant of 4.5. 0.35μm SiGe HBT with an ft of 25 GHz (Toshiba MT4S102T), Si varactor diode with a capacitance ratio of 2.5 : 1 (Toshiba 1SV279), 1005-type chip resistors, and inductors and capacitors are mounted on the substrate by soldering. The circuit size is 16 × 16 × 1.2 mm3.


## 4. Circuit Performance

### 4.1. Frequency-Tunable Performances

Measured gain, input and output return losses for a fixedVC1(CV1) and a variable VC2(CV2) are shown in Figures 7, 8, and 9, respectively. As a coarse tuning, VC2(CV2) was varied from 6 to 18 V with 1 V step since the anode voltage is equal to VCC of 6 V. The maximal reverse voltage is 15 V. f2 moves from 0.73 to 1.31 GHz for a fixed f1 of 0.56 GHz. The gain varied from 0 to 7.2 dB for f2 and keeps a constant value of 8.9 dB for f1. VCC and VC1 were 6 V. Since the LC resonators are used in the output circuit, the output return loss is varied with the resonant frequency, but the input return loss keeps constant. Since the amplifier employed a reflective match approach for low-noise figure, the input and output return losses were less than 10 dB, which can be greatly improved by employing the lossy match approach using resistors [4]. In [4], the input and output return losses can be improved up to greater than 10 dB across the same frequency band by using the lossy match approach. This approach, however, provides a low gain and a poor noise figure. Thus, the reflective match approach was chosen. Meanwhile, the measured gain, input and output return losses for a variable VC1(CV1) and a fixed VC2(CV2) are shown in Figures 10, 11, and 12, respectively. In a similar way, VC1(CV1) was varied from 6 to 18 V with 1 V step. f1 moves from 0.57 to 0.97 GHz and f2 from 1.31 to 1.34 GHz. The gain varied from 8.7 to 13.1 dB for f1 and 1.3 to 6.5 dB for f2. VCC and VC2 were 6 V. If the varactor-loaded LC resonator shows an ideally open-circuited performance, f1 and f2 can be controlled independently. But actually the gain and frequency shift at f2 was affected to a certain extent by controlling VC1(CV1). This is because the parasitic resistance and low-Q factor of the varactor diode (CV1) have made a serious effect on the gain and frequency at f2. Therefore, VC2(CV2) in addition to VC1(CV1) must be adjusted simultaneously for keeping f2 constant. Moreover, as f1 comes close to f2, the bandwidth at f1 becomes wider, and the gain decreases in Figure 10. This is considered to be the same reason of the parasitic resistance and low-Q factor of the varactor diode. In Figures 7 and 10, the 3rd gain peaking can be seen around 1.6 GHz. This is mainly due to the fact that the LC resonator in Figure 1 shows a dual-band resonation due to the lead inductance of the varactor diode.Figure 7
Measured gain for a fixedVC1(CV1) and a variable VC2(CV2).Figure 8
Measured input return loss for a fixedVC1(CV1) and a variable VC2(CV2).Figure 9
Measured output return loss for a fixedVC1(CV1) and a variable VC2(CV2).Figure 10
Measured gain for a fixedVC2(CV2) and a variable VC1(CV1).Figure 11
Measured input return loss for a fixedVC2(CV2) and a variable VC1(CV1).Figure 12
Measured output return loss for a fixedVC2(CV2) and a variable VC1(CV1).


### 4.2. Phase-Shifting Performances

Measured phase variations for a fixedVC1(CV1) and a variable VC2(CV2) are shown in Figure 13. As a fine tuning, VC2(CV2) was varied from 10 to 12 V with 0.2 V step. f2 moves from 0.92 to 1.01 GHz for a fixed f1 of 0.63 GHz. The phase variations at 0.97 GHz and 0.63 GHz were 90.3° and 2.5°, respectively. VCC and VC1 were 6 V and 7 V, respectively. The gain variation, however, was 7 dB at 0.97 GHz. The gain variation can be improved by using a lossy match approach, which in turn degrades a phase-shifting amount; that is, there exists a design tradeoff between phase and gain variations. As one approach to address this problem, an additional variable gain amplifier (VGA) with a constant phase is utilized to compensate for the gain variation. Meanwhile, the measured phase variations for a fixed VC2(CV2) and a variable VC1(CV1) are shown in Figure 14. In a similar way, VC1(CV1) was varied from 6 to 8 V with 0.2 V step. f1 moves from 0.56 to 0.7 GHz for a fixed f2 of 0.97 GHz. The phase variation at 0.63 GHz was 99°. The phase variation at 0.97 GHz, however, was less than 2.1°. VCC and VC2 were 6 V and 11 V, respectively.Figure 13
Measured phase variations for a fixedVC1(CV1) and a variable VC2(CV2).Figure 14
Measured phase variations for a fixedVC2(CV2) and a variable VC1(CV1).


### 4.3. Noise Figure Performances

Noise figure performances of the dual-band frequency-tunable and phase-shifting differential amplifier have been measured. The measured noise figures are plotted in Figure15 for a fixed VC1(CV1) and a variable VC2(CV2) and in Figure 16 for a variable VC1(CV1) and a fixed VC2(CV2). VCC, VC1, and VC2 were 6 V, 7 V and 9–13 V in Figure 15. Meanwhile, VCC,VC1, and VC2 were 6 V, 5–9 V, and 11 V in Figure 16. The measured noise figures were less than 4.1 dB at 0.63 GHz and less than 5.3 dB at 0.97 GHz in Figure 15. Meanwhile, the measured noise figures were better than 4.4 dB at 0.63 GHz and better than 4.3 dB at 0.97 GHz in Figure 16. As compared with Figures 7 and 10, a narrow variable range was chosen for VC1 and VC2 since it becomes difficult to show all noise figure data at the same time and make clear the difference. Overall the noise figure performances are not seriously affected by the frequency tuning.Figure 15
Measured noise figure performances for a fixedVC1(CV1) and a variable VC2(CV2).Figure 16
Measured noise figure performances for a fixedVC2(CV2) and a variable VC1(CV1).


### 4.4. IIP3

IIP3 performances were measured with two tones of a 10 MHz separation for a fixed VC1 (6 V) and a variable VC2 (6–18 V) as well as a variable VC1 (6–18 V) and a fixed VC2 (6 V). VCC was 6 V. The amplifier has shown the minimum IIP3 of −8.7 dBm for a fixed VC1 (6 V) and a variable VC2 (6–18 V) as well as −6.5 dBm for a variable VC1 (6–18 V) and a fixed VC2 (6 V). A total collector current IC appeared in all measurements and ranged from 9 to 12 mA.


## 4.1. Frequency-Tunable Performances

Measured gain, input and output return losses for a fixedVC1(CV1) and a variable VC2(CV2) are shown in Figures 7, 8, and 9, respectively. As a coarse tuning, VC2(CV2) was varied from 6 to 18 V with 1 V step since the anode voltage is equal to VCC of 6 V. The maximal reverse voltage is 15 V. f2 moves from 0.73 to 1.31 GHz for a fixed f1 of 0.56 GHz. The gain varied from 0 to 7.2 dB for f2 and keeps a constant value of 8.9 dB for f1. VCC and VC1 were 6 V. Since the LC resonators are used in the output circuit, the output return loss is varied with the resonant frequency, but the input return loss keeps constant. Since the amplifier employed a reflective match approach for low-noise figure, the input and output return losses were less than 10 dB, which can be greatly improved by employing the lossy match approach using resistors [4]. In [4], the input and output return losses can be improved up to greater than 10 dB across the same frequency band by using the lossy match approach. This approach, however, provides a low gain and a poor noise figure. Thus, the reflective match approach was chosen. Meanwhile, the measured gain, input and output return losses for a variable VC1(CV1) and a fixed VC2(CV2) are shown in Figures 10, 11, and 12, respectively. In a similar way, VC1(CV1) was varied from 6 to 18 V with 1 V step. f1 moves from 0.57 to 0.97 GHz and f2 from 1.31 to 1.34 GHz. The gain varied from 8.7 to 13.1 dB for f1 and 1.3 to 6.5 dB for f2. VCC and VC2 were 6 V. If the varactor-loaded LC resonator shows an ideally open-circuited performance, f1 and f2 can be controlled independently. But actually the gain and frequency shift at f2 was affected to a certain extent by controlling VC1(CV1). This is because the parasitic resistance and low-Q factor of the varactor diode (CV1) have made a serious effect on the gain and frequency at f2. Therefore, VC2(CV2) in addition to VC1(CV1) must be adjusted simultaneously for keeping f2 constant. Moreover, as f1 comes close to f2, the bandwidth at f1 becomes wider, and the gain decreases in Figure 10. This is considered to be the same reason of the parasitic resistance and low-Q factor of the varactor diode. In Figures 7 and 10, the 3rd gain peaking can be seen around 1.6 GHz. This is mainly due to the fact that the LC resonator in Figure 1 shows a dual-band resonation due to the lead inductance of the varactor diode.Figure 7
Measured gain for a fixedVC1(CV1) and a variable VC2(CV2).Figure 8
Measured input return loss for a fixedVC1(CV1) and a variable VC2(CV2).Figure 9
Measured output return loss for a fixedVC1(CV1) and a variable VC2(CV2).Figure 10
Measured gain for a fixedVC2(CV2) and a variable VC1(CV1).Figure 11
Measured input return loss for a fixedVC2(CV2) and a variable VC1(CV1).Figure 12
Measured output return loss for a fixedVC2(CV2) and a variable VC1(CV1).


## 4.2. Phase-Shifting Performances

Measured phase variations for a fixedVC1(CV1) and a variable VC2(CV2) are shown in Figure 13. As a fine tuning, VC2(CV2) was varied from 10 to 12 V with 0.2 V step. f2 moves from 0.92 to 1.01 GHz for a fixed f1 of 0.63 GHz. The phase variations at 0.97 GHz and 0.63 GHz were 90.3° and 2.5°, respectively. VCC and VC1 were 6 V and 7 V, respectively. The gain variation, however, was 7 dB at 0.97 GHz. The gain variation can be improved by using a lossy match approach, which in turn degrades a phase-shifting amount; that is, there exists a design tradeoff between phase and gain variations. As one approach to address this problem, an additional variable gain amplifier (VGA) with a constant phase is utilized to compensate for the gain variation. Meanwhile, the measured phase variations for a fixed VC2(CV2) and a variable VC1(CV1) are shown in Figure 14. In a similar way, VC1(CV1) was varied from 6 to 8 V with 0.2 V step. f1 moves from 0.56 to 0.7 GHz for a fixed f2 of 0.97 GHz. The phase variation at 0.63 GHz was 99°. The phase variation at 0.97 GHz, however, was less than 2.1°. VCC and VC2 were 6 V and 11 V, respectively.Figure 13
Measured phase variations for a fixedVC1(CV1) and a variable VC2(CV2).Figure 14
Measured phase variations for a fixedVC2(CV2) and a variable VC1(CV1).


## 4.3. Noise Figure Performances

Noise figure performances of the dual-band frequency-tunable and phase-shifting differential amplifier have been measured. The measured noise figures are plotted in Figure15 for a fixed VC1(CV1) and a variable VC2(CV2) and in Figure 16 for a variable VC1(CV1) and a fixed VC2(CV2). VCC, VC1, and VC2 were 6 V, 7 V and 9–13 V in Figure 15. Meanwhile, VCC,VC1, and VC2 were 6 V, 5–9 V, and 11 V in Figure 16. The measured noise figures were less than 4.1 dB at 0.63 GHz and less than 5.3 dB at 0.97 GHz in Figure 15. Meanwhile, the measured noise figures were better than 4.4 dB at 0.63 GHz and better than 4.3 dB at 0.97 GHz in Figure 16. As compared with Figures 7 and 10, a narrow variable range was chosen for VC1 and VC2 since it becomes difficult to show all noise figure data at the same time and make clear the difference. Overall the noise figure performances are not seriously affected by the frequency tuning.Figure 15
Measured noise figure performances for a fixedVC1(CV1) and a variable VC2(CV2).Figure 16
Measured noise figure performances for a fixedVC2(CV2) and a variable VC1(CV1).


## 4.4. IIP3

IIP3 performances were measured with two tones of a 10 MHz separation for a fixed VC1 (6 V) and a variable VC2 (6–18 V) as well as a variable VC1 (6–18 V) and a fixed VC2 (6 V). VCC was 6 V. The amplifier has shown the minimum IIP3 of −8.7 dBm for a fixed VC1 (6 V) and a variable VC2 (6–18 V) as well as −6.5 dBm for a variable VC1 (6–18 V) and a fixed VC2 (6 V). A total collector current IC appeared in all measurements and ranged from 9 to 12 mA.


## 5. Conclusions

The dual-band SiGe HBT frequency-tunable and phase-shifting differential amplifier has been presented for use in the future active phased array antennas with a multiband, multibeam, and multi-target tracking operation. To realize multifunction capabilities including frequency-tunable and phase-shifting capabilities, the novel design approach to incorporate varactor-loaded, stacked LC resonators with different resonant frequencies into the circuit design has been proposed. Then to make sure the usefulness of the design approach, the dual-band frequency-tunable and phase-shifting differential amplifier using SiGe HBT and Si varactor diode have been actually designed, fabricated, and tested. It is confirmed from the measured results that the design approach proposed in this paper would be one of the candidates for the multifunctional amplifier for use in the current and future wireless, aerospace, microwave, and millimeter-wave sensors and radar systems.



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*Source: 157971-2012-10-21.xml*