Patent ID: 12218438

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

A plurality of preferred embodiments of the present invention will be described below with reference to specific examples and the drawings. The same reference numerals denote the same or substantially the same portions and elements in the drawings. Considering the explanation of main points and to facilitate understanding, the preferred embodiments are separately illustrated for convenience. However, portions and elements illustrated in different preferred embodiments may be replaced or combined. In second and subsequent preferred embodiments, a description of portions and elements that are common to a first preferred embodiment will be omitted, and only different points will be described. In particular, the same or substantially the same advantageous effects obtained by the same or substantially the same configuration will not be repeated in the preferred embodiments.

The “antenna device” illustrated in the preferred embodiments is applicable to one that transmits signals or one that receives signals. Even in a case in which the “antenna device” is described as an antenna that radiates electromagnetic waves, the antenna device is not limited to a source that generates the electromagnetic waves. Also in a case of receiving an electromagnetic wave that is generated by a communication-partner antenna device, that is, even when the transmission and reception are reversed, the same or substantially the same advantageous effects are produced.

First Preferred Embodiment

FIG.1is a perspective view illustrating a main configuration of an antenna device101according to the first preferred embodiment and electronic equipment including the antenna device101.FIG.2is a plan view of a main portion of the antenna device101.

A metal housing of the electronic equipment includes a radiating element10, which is an end portion of the metal housing, and a metal housing main portion40. The metal housing main portion40includes a planar portion41and side surface portions42and43.

The antenna device101includes the radiating element10, a non-radiating resonant circuit20, and the coupling circuit30.

The radiating element10is defined by the end portion of the metal housing, and includes an end surface portion11and side surface portions12and13. An end portion of the side surface portion12is connected to a ground (is grounded) of a circuit substrate6via an inductor8. Although an end portion of the side surface portion13is open, a parasitic capacitance C is generated between this open end and the ground. Note that a connector7, such as a USB, for example, is mounted on the circuit substrate6, and an opening for the connector7is provided in the end surface portion11. However, the connector7is not a component of the antenna device101.

The circuit substrate6includes a ground region GZ in which a ground electrode GND is provided and a non-ground region NGZ in which a ground electrode is not provided. The end portion of the metal housing, which defines the radiating element10, is located on the non-ground region side. In the non-ground region NGZ of the circuit substrate6, the non-radiating resonant circuit20is defined by a conductor pattern. Also in the non-ground region NGZ of the circuit substrate6, a feeding line9that connects the coupling circuit30and the radiating element10to each other is provided.

As illustrated inFIG.2, the non-radiating resonant circuit20is defined by a linear conductor pattern including a returning portion20FB along the linear conductor pattern. In this manner, since the linear conductor pattern including a returning portion along the linear conductor pattern is provided, the non-radiating resonant circuit20is provided in a small area, and an electric length that is necessary for resonance is able to be obtained. In addition, in the present preferred embodiment, a first linear conductor pattern portion21extending from the coupling circuit30and a second linear conductor pattern portion22that returns to a side away from the radiating element10are included. With this structure, since a portion close to the radiating element10(the end surface portion11in particular) is short, and extending directions are opposite to each other, substantial coupling with the radiating element10(the end surface portion11in particular) is weak. This reduces or prevents unnecessary coupling between the non-radiating resonant circuit20and the radiating element10.

Note that the second linear conductor pattern portion22is preferably wider than the first linear conductor pattern portion21. Thus, a resonant bandwidth is able to be broadened.

FIG.3is a plan view illustrating a position at which the non-radiating resonant circuit20is provided. The radiating element10is defined by a conductive member (the end surface portion11and the side surface portions12and13) that includes three sides in a plan view, and the non-radiating resonant circuit is surrounded by a radiating element formation region10Z surrounded by the three sides of the radiating element10in a plan view. The entire or substantially the entire non-radiating resonant circuit20does not have to be provided within the radiating element formation region10Z, and a about half or more of the non-radiating resonant circuit20is preferably included within the radiating element formation region10Z. Since the non-radiating resonant circuit20is not used as a radiating element, the non-radiating resonant circuit20is preferably “non-radiating”. Thus, in a case in which the non-radiating resonant circuit20is surrounded by the three sides of the conductive member in a plan view and about half or more of the non-radiating resonant circuit20is included within the radiating element formation region10Z, the non-radiating resonant circuit20is shielded by the radiating element10. This increases a non-radiating property of the non-radiating resonant circuit20seen from a distance.

FIG.4illustrates a configuration of the coupling circuit30and a circuit connected thereto. The coupling circuit includes a first coupling element31and a second coupling element32that is coupled to the first coupling element31, and a transformer is defined by the first coupling element31and the second coupling element32. The first coupling element31and the second coupling element32have small inductances, each of which is preferably, for example, about 10 nH or less. The radiating element10and the non-radiating resonant circuit20are coupled to each other via the coupling circuit30preferably with a coupling coefficient of, for example about 0.5 or more, and more preferably with a coupling coefficient of, for example, about 0.8 or more. When the inductance of a coupling element is smaller, an influence on a circuit characteristic and a radiation characteristic of the radiating element10is able to be further reduced or prevented. When the coupling coefficient is higher, the radiating element10and the non-radiating resonant circuit20are able to be more electrically connected to each other, and a resonance point is able to be added to only a frequency at which the non-radiating resonant circuit20more largely contributes to resonance. In this manner, by configuring a transformer in which electromagnetic field coupling is produced between the first coupling element31and the second coupling element32, a coupling circuit with a high coupling coefficient between the first coupling element31and the second coupling element32is provided, and a resonance characteristic of the non-radiating resonant circuit20when viewing the radiating element10from a feeder circuit1is likely to be shown.

The first coupling element31is connected between the radiating element10and the feeder circuit1. A first end of the second coupling element32is connected to the non-radiating resonant circuit20, and a second end thereof is connected to the ground (is grounded) of the circuit substrate6.

With the electronic equipment according to the present preferred embodiment, the metal portion of the housing that accommodates the feeder circuit is used as the radiating element, and thus, it is unnecessary to provide a conductive member or a conductor pattern dedicated to the radiating element, thus reducing the size of the electronic equipment. In addition, also in electronic equipment including a metal housing, the metal housing does not block electromagnetic waves.

FIG.5Ais an equivalent circuit diagram of the antenna device101in a high band. In a high band (e.g., 1.6 GHz to 2.3 GHz), the inductor8(seeFIG.2andFIG.4) has a predetermined high impedance, and a tip of the radiating element10is equivalently open. In this state, the radiating element10serves as a monopole antenna that resonates at ¾ wavelengths or (2n+1)/4 wavelengths (n is a natural number).

FIG.5Bis an equivalent circuit diagram of the antenna device101in a low band. In a low band (e.g., about 700 MHz to about 900 MHz), the inductor8has a predetermined inductance, and the tip of the radiating element10is grounded via the inductor8. In this state, the radiating element10defines and functions as a loop antenna of one wavelength or an integer multiple thereof.

A series circuit including an inductor L20and a capacitor C20illustrated inFIGS.5A and5Bis an element that represents an equivalent circuit in which the non-radiating resonant circuit20is simply illustrated as a lumped constant circuit. The non-radiating resonant circuit20defines and functions as an open stub that resonates at a predetermined frequency at about ¾ wavelengths or about (2n+1)/4 wavelengths (n is a natural number), for example. Thus, the inductor L20and the capacitor C20are used inFIGS.5A and5B. The non-radiating resonant circuit20resonates in, for example, a frequency band whose center is about 2.1 GHz, for example. Note that in the present preferred embodiment, since the non-radiating resonant circuit20has a shape in which the linear conductor pattern is returned, a standing wave is not provided in the linear conductor pattern compared with a simple straight line conductor pattern, and a Q value of resonance as a resonance circuit is relatively small.

FIG.6illustrates a frequency characteristic of a return loss of the antenna device101and an antenna device of a comparative example. InFIG.6, a return loss characteristic RL1is a return loss of the antenna device101according to the present preferred embodiment, and a return loss characteristic RL2is a return loss of the antenna device according to the comparative example. The antenna device according to the comparative example is an antenna device in which the coupling circuit30and the non-radiating resonant circuit20are not included. In either antenna device, a pole is generated at a center frequency F1of a low band (e.g., about 700 MHz to about 900 MHz). This is due to the resonance characteristic of the loop antenna illustrated inFIG.5B. Another pole is generated at a frequency F2(e.g., around 1.75 GHz). This is due to ¾ wavelength resonance of the monopole antenna illustrated inFIG.5A. Another pole is generated at a frequency F3(e.g., around 2.3 GHz). This is due to 5/4 wavelength resonance of the monopole antenna illustrated inFIG.5A.

Note that it is preferable that a length “r1” of a line between the first coupling element31and the feeder circuit1illustrated inFIG.4and a length “r2” of a line between an end portion of the second coupling element32and the ground are less than about ⅛ wavelength of the resonant frequency, for example. The wavelength here may mean an effective wavelength considering a wavelength shortening effect of a magnetic body or a dielectric. The threshold is set to about “⅛ wavelength” because it is practical until a condition at which ⅛ wavelength current becomes 1/√2, in other words, a power that is able to be transmitted is approximately halved, is satisfied.

Here,FIG.7illustrates a conceptual diagram of a difference in impedance matching depending on the strength of the coupling. InFIG.7, loci T0, T1, and T2are impedance loci representing, on a Smith chart, impedances when seeing the antenna device101from the feeder circuit1. Locus T0is a characteristic in a state in which the coupling circuit30and the non-radiating resonant circuit20are not provided, locus T1is a characteristic in a state in which the first coupling element31and the second coupling element32of the coupling circuit30are appropriately coupled to each other, and locus T2is a characteristic in a state in which the coupling between the first coupling element31and the second coupling element32of the coupling circuit30is too strong.

In this manner, when the coupling between the first coupling element31and the second coupling element32of the coupling circuit30is too strong, the input impedance seen from the feeder circuit deviates from the impedance (e.g., about 50Ω) on the feeder circuit (and transmission line) side. Therefore, it is important that the first coupling element31and the second coupling element32of the coupling circuit30are appropriately coupled to each other. The length “r1” of the line between the first coupling element31and the feeder circuit1and the length “r2” of the line between the end portion of the second coupling element32and the ground are set within a range of less than about ⅛ wavelength of the resonant frequency, and thus, the coupling by the coupling circuit30is able to be appropriately set.

In the antenna device101according to the present preferred embodiment, another pole is generated at a frequency F0(e.g., around 2.1 GHz). This is due to the resonance characteristic of the non-radiating resonant circuit20. That is, since the non-radiating resonant circuit20resonates in a frequency band whose center frequency is about 2.1 GHz, for example, the pole is generated at about 2.1 GHz in the frequency characteristic of a return loss of the antenna device101seen from the feeder circuit1. With the antenna device101according to the present preferred embodiment, a high-band application frequency band is broadened from about 1.6 GHz to about 2.3 GHz.

In the low band, the non-radiating resonant circuit20does not resonate, and the return loss characteristic in the low band is not influenced. That is, the non-radiating resonant circuit20influences the return loss characteristic seen from the feeder circuit1in, for example, a frequency band of about 1.6 GHz or higher, and the non-radiating resonant circuit20has substantially no influence in a frequency band lower than that.

The return loss characteristic at around the frequency F0is determined by the resonance characteristic of the non-radiating resonant circuit20, and accordingly, the return loss characteristic at about the frequency F0can be determined as appropriate by the shape of the conductor pattern that constitutes the non-radiating resonant circuit. In the present preferred embodiment, since the non-radiating resonant circuit20is defined by the linear conductor pattern that includes a returning portion along the linear conductor pattern, the sharpness of resonance of the non-radiating resonant circuit20is degraded, and the non-radiating resonant circuit20is able to attenuate a reflection coefficient in a wide band including the band in which the pole generated at the frequency F0and its periphery.

Note that the non-radiating resonant circuit20that defines and functions as an open stub is provided independently or substantially independently of the radiating element10. Thus, there is no influence on a low band, unlike in a case in which a stub is provided in the radiating element, for example.

Next, a configuration of the coupling circuit30will be described.FIG.8is a perspective view of the coupling circuit30, andFIG.9is an exploded plan view illustrating conductor patterns provided on layers of the coupling circuit.

The coupling circuit30included in the antenna device according to the present preferred embodiment is preferably, for example, a rectangular or substantially rectangular parallelepiped chip component to be mounted on the circuit substrate6. InFIG.8, an external configuration of the coupling circuit30and an internal structure thereof are separately illustrated. The external configuration of the coupling circuit30is represented by a two-dotted-and-dashed line. On an outer surface of the coupling circuit30, a feeder circuit connection terminal PF, a radiating element connection terminal PA, a ground terminal PG, and a non-radiating resonant circuit connection terminal PS are provided. In addition, the coupling circuit30includes a first surface MS1and a second surface MS2that is opposed to the first face. In the present preferred embodiment, the first surface MS1is a mount surface, and this surface faces a circuit substrate. On a top surface (second surface) that is opposed to the mount surface (first surface) MS1, a direction discrimination mark DDM is provided. This direction discrimination mark DDM is used to detect a direction of a chip component when, for example, the coupling circuit30is mounted as the chip component on a circuit substrate by a mounter.

Inside the coupling circuit30, a first conductor pattern L11, a second conductor pattern L12, a third conductor pattern L21, and a fourth conductor pattern L22are provided. The first conductor pattern L11and the second conductor pattern L12are connected to each other by an interlayer connection conductor V1. The third conductor pattern L21and the fourth conductor pattern L22are connected to each other by an interlayer connection conductor V2. Note thatFIG.8illustrates insulating materials S11, S12, S21, and S22, on which the respective conductor patterns are provided, separately in a stacking direction. These insulating materials S11, S12, S21, and S22may preferably be a non-magnetic ceramic multi-layer body made of, for example, LTCC (Low Temperature Co-fired Ceramics) or other suitable material, or may be a resin multi-layer body preferably made of, for example, a resin material, such as polyimide or liquid crystal polymer. In this manner, since the material layers are non-magnetic (not a magnetic ferrite), it is possible to use the material layers for a coupling circuit even in a high frequency band exceeding several hundreds of MHz.

Each of the conductor patterns and the interlayer connection conductors is preferably made of, for example, a conductor material including Ag or Cu as a main component and having a small resistivity. In a case in which the material layers are ceramic, for example, the conductor patterns and the interlayer connection conductors are formed by screen printing and firing of a conductive paste including Ag or Cu as a main component. In a case in which the material layers are resin, for example, the conductor patterns and the interlayer connection conductors are patterned by etching or other suitable method of a metal foil, such as an Al foil or a Cu foil, for example.

As illustrated inFIG.9, the first conductor pattern L11, the second conductor pattern L12, the third conductor pattern L21, and the fourth conductor pattern L22are provided in this order from a layer close to the mount surface. A first end of the first conductor pattern L11is connected to the radiating element connection terminal PA, and a second end thereof is connected to a first end of the second conductor pattern L12via the interlayer connection conductor V1. A second end of the second conductor pattern L12is connected to the feeder circuit connection terminal PF. In addition, a first end of the third conductor pattern L21is connected to a non-radiating resonant circuit connection terminal PS, and a second end of the third conductor pattern L21is connected to a first end of the fourth conductor pattern L22via the interlayer connection conductor V2. A second end of the fourth conductor pattern L22is connected to the ground terminal PG.

In addition, a winding direction of the first coupling element31from the feeder circuit connection terminal PF to the radiating element connection terminal PA and a winding direction of the second coupling element32from the non-radiating resonant circuit connection terminal PS to the ground terminal PG are opposite to each other. That is, a magnetic field (magnetic flux) generated when current flows in the first coupling element31from the feeder circuit connection terminal PF to the radiating element connection terminal PA and a magnetic field (magnetic flux) generated when current flows in the second coupling element32from the non-radiating resonant circuit connection terminal PS to the ground terminal PG weaken each other. Here, when the radiating element connection terminal PA resonates as a monopole antenna, the first coupling element31and the second coupling element32have opposite polarities in the coupling circuit30that is connected via the feeder circuit1and the ground electrode GND. Current flows in the first coupling element31from the feeder circuit connection terminal PF to the radiating element connection terminal PA, and current flows in the second coupling element32from the non-radiating resonant circuit connection terminal PS to the ground terminal PG. Magnetic fields (magnetic fluxes) that are generated weaken each other. Thus, a mutual inductance due to the coupling between the first coupling element31and the second coupling element32decreases the inductances of the first coupling element31and the second coupling element32, so as to have little influence on the circuit characteristic and the radiation characteristic of the radiating element10.

FIG.10is a circuit diagram of the coupling circuit30including the four coil conductor patterns. The second conductor pattern L12and the first conductor pattern L11are connected in series to define the first coupling element31. Similarly, the fourth conductor pattern L22and the third conductor pattern L21are connected in series to define the second coupling element32. The second conductor pattern L12and the third conductor pattern L21are adjacent to each other in the thickness direction, and the magnetic field coupling between the second conductor pattern L12and the third conductor pattern L21is particularly strong. Thus, the second conductor pattern L12and the third conductor pattern L21are adjacent to each other inFIG.10. Obviously, magnetic field coupling is established between the second conductor pattern L12and the fourth conductor pattern L22and between the first conductor pattern L11and the third conductor pattern L21.

In the example illustrated inFIG.9, a capacitor formation conductor pattern C11is provided in a portion of the second conductor pattern L12, and a capacitor formation conductor pattern C12is provided in a portion of the third conductor pattern L21. Accordingly, as illustrated inFIG.10, a capacitor C1is provided between a middle of the second conductor pattern L12and the non-radiating resonant circuit connection terminal PS. The capacitor C1defines and functions as an impedance matching circuit between the feeder circuit1and the non-radiating resonant circuit20.

Second Preferred Embodiment

FIG.11illustrates a circuit configuration of an antenna device102according to the second preferred embodiment of the present invention. In the antenna device102, an inductor35is connected (inserted) between the second coupling element32of the coupling circuit30and the non-radiating resonant circuit20. The remaining configuration is the same as or similar to that of the circuit illustrated inFIG.4in the first preferred embodiment.

According to the present preferred embodiment, since the inductor35is provided to a portion at which current is low, while a change in the coupling of the coupling circuit30is reduced or prevented, the resonant frequency of the non-radiating resonant circuit20is able to be decreased, and a desired communication band is able to be obtained. Alternatively, while the resonant frequency is maintained, the length of the non-radiating resonant circuit20is able to be reduced, and thus the area used is able to be reduced.

Note that the inductor35may also be integrated with the coupling circuit30. However, it is preferable that the inductor35is not coupled to the first coupling element31.

Third Preferred Embodiment

FIG.12illustrates a circuit configuration of an antenna device103according to a third preferred embodiment of the present invention. In the antenna device103, the inductor35is connected (inserted) between the second coupling element32of the coupling circuit30and the ground. The remaining configuration is the same as or similar to that of the circuit illustrated inFIG.4in the first preferred embodiment.

When the coupling circuit30is added to the antenna device, a parasitic capacitance is generated between the ground and the coupling circuit30. According to the present preferred embodiment, resonance between the inductor35and the parasitic capacitance reduce or prevent a reactance component. Therefore, in a frequency band in which an antenna characteristic is not to be changed by addition of the coupling circuit30to the antenna device, by providing the inductor35with such an inductance as to resonate with the parasitic capacitance, a change from a matching state in which the coupling circuit30is not mounted is able to be reduced or prevented.

In addition, the inclusion of the inductor35is able to decrease the resonant frequency of the non-radiating resonant circuit20, and a desired communication band or communication characteristic is able to be obtained. Alternatively, while the resonant frequency is maintained, the length of the antenna is able to be reduced, and the area used is able to be reduced.

Note that the inductor35may also be integrated with the coupling circuit30. However, it is preferable that the inductor35is not coupled to the first coupling element31.

Fourth Preferred Embodiment

FIG.13illustrates a circuit configuration of an antenna device104according to a fourth preferred embodiment of the present invention. In the antenna device104, a capacitor36is connected (inserted) between the second coupling element32of the coupling circuit30and the non-radiating resonant circuit20. The remaining configuration is the same as or similar to that of the circuit illustrated inFIG.4in the first preferred embodiment.

According to the present preferred embodiment, the resonant frequency on the non-radiating resonant circuit side is able to be increased, and a desired communication band is able to be obtained.

Note that the capacitor36may be integrated with the coupling circuit30.

Fifth Preferred Embodiment

FIG.14illustrates a circuit configuration of an antenna device105according to a fifth preferred embodiment of the present invention. In the antenna device105, the capacitor36is connected (inserted) between the second coupling element32of the coupling circuit30and the ground. The remaining configuration is the same as or similar to that of the circuit illustrated inFIG.4in the first preferred embodiment.

According to the present preferred embodiment, a parasitic capacitance generated between the ground and the coupling circuit30by inclusion of the coupling circuit30is able to be reduced (combined capacitance is able to be reduced), and a change from a matching state in which the coupling circuit30is not provided is able to be reduced or prevented. In addition, the resonant frequency of the non-radiating resonant circuit20is able to be increased, and a desired communication band or communication characteristic is able to be obtained.

Note that the capacitor36may be integrated with the coupling circuit30.

Sixth Preferred Embodiment

FIG.15illustrates a circuit configuration of an antenna device106A according to a sixth preferred embodiment of the present invention. In the antenna device106A, the inductor35is connected (inserted) between the first coupling element31of the coupling circuit30and the radiating element10. The remaining configuration is the same as or similar to that of the circuit illustrated inFIG.4in the first preferred embodiment.

With the configuration of the antenna device106A, the first coupling element31is closer to the feeder circuit1, at which the current is strong, than the inductor35is. Thus, while a power ratio to be supplied to the non-radiating resonant circuit20is maintained, the resonant frequency of the radiating element10is able to be changed, and a level of impedance matching is able to be adjusted. In addition, a self-resonant frequency that is determined by the inductances of the first coupling element31and the second coupling element32and the parasitic capacitance generated between the first coupling element31and the second coupling element32is unlikely to be decreased, and thus, the self-resonant frequency does not adversely affect the use in a communication frequency band. That is, in a state of self-resonance, energy in the frequency band falls to the ground and is not radiated. However, in a state in which the self-resonant frequency is higher than the communication frequency band, such a problem does not arise.

FIG.16illustrates a circuit configuration of an antenna device106B according to the sixth preferred embodiment. In the antenna device106B, the inductor35is connected (inserted) between the first coupling element31of the coupling circuit30and the feeder circuit1. The remaining configuration is the same as or similar to that of the circuit illustrated inFIG.4in the first preferred embodiment.

With the configuration of the antenna device106B, since the first coupling element31of the coupling circuit30is disposed to a side at which current is weaker than that at the position of the inductor35, compared with a case in which the inductor is disposed between the radiating element10and the first coupling element31, it is possible to adjust the level of impedance matching as appropriate in resonance (resonant frequency) added by the coupling circuit30and the non-radiating resonant circuit20. Specifically, it is possible to avoid a situation in which an input impedance excessively changes and the impedance is no longer matched.

In addition, the inclusion of the inductor35is able to decrease the self-resonant frequency of the coupling circuit30, and thus, by setting the self-resonant frequency to a frequency band that is not desired to be radiated, unnecessary radiation is able to be reduced or prevented.

FIG.17illustrates a circuit configuration of an antenna device106C according to the sixth preferred embodiment. In the antenna device106C, the capacitor36is connected (inserted) between the first coupling element31of the coupling circuit30and the radiating element10. The remaining configuration is the same as or similar to that of the circuit illustrated inFIG.4in the first preferred embodiment.

With the configuration of the antenna device106C, by the capacitance of the capacitor36, the resonant frequency of the radiating element10is able to be adjusted, and the level of impedance matching is able to be adjusted.

FIG.18illustrates a circuit configuration of an antenna device106D according to the sixth preferred embodiment. In the antenna device106D, the capacitor36is connected (inserted) between the first coupling element31of the coupling circuit30and the feeder circuit1. The remaining configuration is the same as or similar to that of the circuit illustrated inFIG.4in the first preferred embodiment.

With the configuration of the antenna device106D, by the capacitance of the capacitor36, the resonant frequency of the radiating element10is able to be adjusted, and the level of impedance matching is able to be adjusted. In addition, since the capacitor36is disposed between the feeder circuit1and the first coupling element31, a parasitic capacitance generated between the first coupling element31and the second coupling element32and the capacitor36are connected in series. Accordingly, a combined capacitance included in a self-resonant circuit system is decreased, and the self-resonant frequency is increased. Thus, the self-resonant frequency is able to be excluded from the communication band to be used.

Seventh Preferred Embodiment

FIG.19Aillustrates a circuit configuration of an antenna device107A according to a seventh preferred embodiment of the present invention, andFIG.19Billustrates a circuit configuration of an antenna device107B according to the seventh preferred embodiment. The configuration of these antenna devices107A and107B is the same as or similar to that of the circuit illustrated inFIG.4in the first preferred embodiment. However, when a self-inductance of the first coupling element31of the coupling circuit30is represented as L1and a self-inductance of the second coupling element32is represented as L2, the first coupling element31and the second coupling element32of the coupling circuit30preferably have a relationship of L2>L1in the antenna device107A, and a relationship of L2<L1in the antenna device107B. With the relationship of L2>L1, compared with a case in which L1=L2, the resonant frequency of the non-radiating resonant circuit20is able to be decreased. Alternatively, when comparison is made at the same resonant frequency, the non-radiating resonant circuit20is able to be shortened.

In addition, when L2>L1, compared with a configuration in which the inductor is connected (added) to the second coupling element32outside the coupling circuit30, the entire or substantially the entire second coupling element32with a relatively large self-inductance contributes to the coupling with the first coupling element31. Thus, a power ratio to be supplied to the non-radiating resonant circuit20is able to be increased.

In addition, when L2<L1, compared with a configuration in which the inductor is connected (added) to the first coupling element31outside the coupling circuit30, the entire or substantially the entire first coupling element31with a relatively large self-inductance contributes to the coupling with the second coupling element32. Thus, a power ratio to be supplied to the non-radiating resonant circuit20is able to be increased.

FIG.20is an exploded plan view illustrating conductor patterns provided on layers of the coupling circuit30according to the present preferred embodiment. The coupling circuit30included in an antenna device according to the present preferred embodiment is preferably, for example, a rectangular or substantially rectangular parallelepiped chip component to be mounted on a circuit substrate.

On insulating materials S11, S12, S21, S22, and S23, conductor patterns L11, L12, L21, L22, and L23are respectively provided. A first end of the conductor pattern L11is connected to the radiating element connection terminal PA, and a second end thereof is connected to a first end of the conductor pattern L12via the interlayer connection conductor V1. A second end of the conductor pattern L12is connected to the feeder circuit connection terminal PF. A first end of the conductor pattern L21is connected to the non-radiating resonant circuit connection terminal PS, and a second end thereof is connected to a first end of the conductor pattern L22via an interlayer connection conductor V21. A second end of the conductor pattern L22is connected to a first end of the conductor pattern L23via an interlayer connection conductor V22. A second end of the conductor pattern L23is connected to the ground terminal PG.

FIG.21is a sectional view of the coupling circuit30.FIG.22is a plan view illustrating overlap between the conductor pattern L12and the conductor pattern L21in particular. A coil opening or a coil diameter of the conductor patterns L11and L12of the first coupling element31is smaller than a coil opening or a coil diameter of the conductor patterns L21, L22, and L23of the second coupling element32. In addition, portions of the conductor patterns L11and L12and the conductor patterns L21, L22, and L23overlap with each other. In the example illustrated inFIG.21andFIG.22, about ½ of the width of the conductor patterns is overlapped along the entire or substantially the entire circumference.

FIG.23is an exploded plan view illustrating conductor patterns provided on layers of another coupling circuit30according to the seventh preferred embodiment. The shape and size of the conductor patterns differ from those in the example illustrated inFIG.20. Among the conductor patterns of the coupling circuit illustrated inFIG.23, a coil outer diameter of the conductor patterns L11and L12of the first coupling element31is smaller than a coil inner diameter of the conductor patterns L21, L22, and L23of the second coupling element32.

With the configuration illustrated inFIG.20toFIG.23, a parasitic capacitance generated between the conductor patterns (L11and L12) of the first coupling element31and the conductor patterns (L21, L22, and L23) of the second coupling element32is reduced or prevented. Accordingly, the self-resonant frequency determined by the inductances of the first coupling element31and the second coupling element32and the above parasitic capacitance is increased, and the self-resonant frequency is able to be excluded from the communication band to be used. In addition, even if the conductor patterns (L11and L12) of the first coupling element31and the conductor patterns (L21, L22, and L23) of the second coupling element32are misaligned in a plane direction (X-Y plane direction illustrated inFIG.22), the portion at which the coil opening of the first coupling element31and the coil opening of the second coupling element32overlap with each other is consistently maintained. Accordingly, only a small change in the coupling degree of magnetic field coupling between the first coupling element31and the second coupling element32is produced by plane-direction misalignment of the conductor patterns (L11and L12) constituting the first coupling element31and the conductor patterns (L21, L22, and L23) constituting the second coupling element32.

The examples inFIG.20andFIG.23are both examples of a coupling circuit in which the relationship L1<L2is satisfied. When L1>L2, the first coupling element31may be defined by conductor patterns having a relatively large coil opening.

Note thatFIG.20andFIG.23illustrate examples in which an influence due to misalignment of the conductor patterns (L11and L12) of the first coupling element31and the conductor patterns (L21, L22, and L23) of the second coupling element32is reduced. Similarly, an influence due to plane-direction misalignment of the conductor patterns of the first coupling element31and an influence due to plane-direction misalignment of the conductor patterns of the second coupling element32is able to be reduced. For example, coil openings or coil diameters of the conductor patterns L11and L12that are adjacent to each other in the stacking direction may differ from each other, and portions of line width thereof may overlap with each other in the structure. Similarly, for example, coil openings or coil diameters of the conductor patterns L21, L22, and L23that are adjacent to one another in the stacking direction may differ from one another, and portions of line width thereof may overlap with one another in the structure.

Eighth Preferred Embodiment

FIG.24illustrates a circuit configuration of an antenna device108according to an eighth preferred embodiment of the present invention. The antenna device108includes a first coupling circuit30A, a second coupling circuit30B, a first non-radiating resonant circuit20A, and a second non-radiating resonant circuit20B. The second coupling circuit30B includes a third coupling element33and a fourth coupling element34that are coupled to each other. The third coupling element33of the second coupling circuit30B is connected between the first coupling element31and the feeder circuit1. The first non-radiating resonant circuit20A is connected to the second coupling element32, and the second non-radiating resonant circuit20B is connected to the fourth coupling element34. The remaining configuration is the same as or similar to that of the circuit illustrated inFIG.4in the first preferred embodiment.

A resonant frequency of the first non-radiating resonant circuit20A and a resonant frequency of the second non-radiating resonant circuit20B differ from each other, and thus, a plurality of poles in accordance with these resonant frequencies are generated, and a communication bandwidth is broadened. In addition, if the resonant frequency of the first non-radiating resonant circuit20A and the resonant frequency of the second non-radiating resonant circuit20B are equal or substantially equal to each other, the poles generated in the two non-radiating resonant circuits become deeper, and impedance matching in this frequency band is improved.

Ninth Preferred Embodiment

FIG.25illustrates a circuit configuration of an antenna device109according to a ninth preferred embodiment of the present invention. The antenna device109includes the first coupling circuit30A, the second coupling circuit30B, the first non-radiating resonant circuit20A, and the second non-radiating resonant circuit20B. The second coupling circuit30B includes the third coupling element33and the fourth coupling element34that are coupled to each other.

The third coupling element33is connected between the second coupling element32and the first non-radiating resonant circuit20A. The first non-radiating resonant circuit20A is connected to the second coupling element32, and the second non-radiating resonant circuit20B is connected to the fourth coupling element34. The remaining configuration is the same as or similar to that of the circuit illustrated inFIG.4in the first preferred embodiment.

In the present preferred embodiment, the resonant frequency of the first non-radiating resonant circuit20A and the resonant frequency of the second non-radiating resonant circuit20B are equal or substantially equal to each other, and thus, the poles generated in the two non-radiating resonant circuits become deeper, and impedance matching in this frequency band is improved.

Tenth Preferred Embodiment

FIG.26is a circuit diagram of an antenna device110according to a tenth preferred embodiment of the present invention. The antenna device110includes a switch37connected between the non-radiating resonant circuit20and the ground. The antenna device110also includes a switch38connected between the radiating element10and the ground. The remaining configuration is the same as or similar to that of the circuit illustrated inFIG.4in the first preferred embodiment.

The switches37and38are switched independently or in association with each other. By changing the frequency of a pole generated by providing the coupling circuit30and the non-radiating resonant circuit20in accordance with the state of the switch37, or by changing a matching state, the impedance matching is able to be improved. In addition, by changing the resonant frequency of the non-radiating resonant circuit20or by changing the impedance matching state between the coupling circuit30and the non-radiating resonant circuit20so as to make the non-radiating resonant circuit20be coupled easily to the feeder circuit1via the coupling circuit30, the impedance matching is able to be improved.

In addition, in accordance with the state of the switch38, the frequency of a pole generated by resonance of the radiating element10is able to be changed.

Eleventh Preferred Embodiment

FIGS.27A and27Billustrate circuit configurations of antenna devices111A and111B according to an eleventh preferred embodiment of the present invention. In the examples in both ofFIGS.27A and27B, parasitic capacitances represented by capacitors Cs1and Cs2, for example, are included between the first coupling element31and the second coupling element32of the coupling circuit30. In addition, the coupling circuit30includes an inductor L3connected between the first coupling element31and the second coupling element32.

The inductor L3and the capacitors Cs1and Cs2of parasitic capacitances resonate in parallel. Accordingly, a reactance component generated in the coupling circuit30is reduced or prevented in the parallel resonant frequency band. Thus, a reactance component that is added by including the coupling circuit30is canceled, and a change from a matching state in which the coupling circuit30is not provided is able to be reduced or prevented.

Twelfth Preferred Embodiment

FIG.28illustrates a circuit configuration of an antenna device112according to a twelfth preferred embodiment of the present invention.

The antenna device112according to the present preferred embodiment includes the radiating element10, the coupling circuit30, and the non-radiating resonant circuit20. The feeder circuit1is connected to the radiating element10. The coupling circuit30includes the first coupling element31that is connected between the radiating element10and the ground, and the second coupling element32coupled to the first coupling element31. The non-radiating resonant circuit20is connected to the second coupling element32. Also, the inductor35is disposed between the first coupling element31and the ground in this example.

With the above configuration, the radiating element10and the non-radiating resonant circuit20do not interfere with each other regarding radiation, and a radiation characteristic of the radiating element10is not adversely affected. In addition, a frequency characteristic of a return loss of the radiating element10seen from the feeder circuit1is adjusted by a resonance characteristic of the non-radiating resonant circuit20, and a pole is generated in a desired frequency band to broaden the band of the frequency characteristic of the antenna. Since a current intensity is particularly high in a portion that is connected to the ground, the radiating element10and the non-radiating resonant circuit20are able to be coupled to each other via the coupling circuit30. In addition, the coupling circuit30and the non-radiating resonant circuit20are able to be arranged with a higher degree of freedom.

Thirteenth Preferred Embodiment

FIG.29illustrates a circuit configuration of an antenna device113according to a thirteenth preferred embodiment of the present invention.

The antenna device113according to the present preferred embodiment includes a substrate5on which the coupling circuit30and the non-radiating resonant circuit20are each provided using conductor patterns. The remaining configuration is the same as or similar to that of the circuit illustrated in FIG.4in the first preferred embodiment.

The substrate5is preferably made of, for example, a resin multi-layer substrate or a ceramic multi-layer substrate. In a case of a resin multi-layer substrate, for example, a plurality of thermoplastic resin materials on surfaces of which copper-foil patterns are provided are stacked and pressed with heat. In a case of a ceramic multi-layer substrate, a plurality of ceramic green sheets on surfaces of which conductor-paste patterns are provided are stacked and fired.

Note that in a case in which the coupling circuit30and the non-radiating resonant circuit20are provided on different substrates, the non-radiating resonant circuit20may be provided using the resin multi-layer substrate or the ceramic multi-layer substrate.

According to the present preferred embodiment, since the coupling circuit30and the non-radiating resonant circuit20are integrated with each other, the area used is reduced.

Fourteenth Preferred Embodiment

A fourteenth preferred embodiment of the present invention will illustrate an antenna device including a PIFA (planar inverted-F antenna) and a parasitic radiating element.

FIG.30illustrates a circuit configuration of an antenna device114according to the fourteenth preferred embodiment. The antenna device114according to the present preferred embodiment includes a feeding radiating element10A, a feeding line10AF, a parasitic radiating element10B, and the coupling circuit30. The feeder circuit1is connected between the feeding line10AF and the ground. The configuration and advantageous effects of the coupling circuit30are as described in the above-described preferred embodiments.

The first coupling element31of the coupling circuit30is connected between a connection point Ps between the feeding radiating element10A and the feeding line AF and the ground. The feeding radiating element10A, the feeding line10AF, and the first coupling element31define a PIFA. That is, the first coupling element31of the coupling circuit30is provided at a portion of a short pin of the PIFA. The short pin connects the connection point Ps and the ground to each other. A capacitor or an inductor may be provided in this portion.

The parasitic radiating element10B is preferably a monopole parasitic radiating element, for example. The second coupling element32of the coupling circuit30is disposed in the vicinity of a ground end of the parasitic radiating element10B.

A resonant current iA of the feeding radiating element flows between an open end of the feeding radiating element10A and a ground end of the first coupling element31. In addition, a resonant current iB flows between an open end of the parasitic radiating element10B and a ground end of the second coupling element32. A phase of the current iA flowing in the feeding radiating element10A and a phase of the current iB flowing in the parasitic radiating element10B are different from each other.

In general, if the phase of resonance of the feeding radiating element and the phase of resonance of the parasitic radiating element are the same, a notch is present between the two resonant frequencies in a frequency characteristic of the antenna device. Therefore, the band cannot be broadened even if the parasitic radiating element is provided. That is, the parasitic radiating element cannot be provided adjacent to the feeding radiating element in order to broaden the band.

In contrast, in the present preferred embodiment, the current flowing in the first coupling element31of the coupling circuit30and the current flowing in the second coupling element32have a phase difference. Therefore, the phase of resonance of the feeding radiating element10A and the phase of resonance of the parasitic radiating element10B are not the same, and thus, a notch is not present between the two resonant frequencies. The phase difference between the first coupling element31and the second coupling element32is preferably, for example about 180° at most, and a phase difference of less than or equal to about 180° is generated by a parasitic component. That is, due to an effect of the parasitic capacitance between the first coupling element31and the second coupling element32, the phase difference between the current flowing in the first coupling element31and the current flowing in the second coupling element32is preferably greater than about 0° and less than about 180°, for example.

As illustrated inFIG.30, since the resonant current iA flows between the open end and the short position in the PIFA, the phase of current flowing in the feeder circuit1and the phase of the resonant current iA are different from each other. Accordingly, if the first coupling element31of the coupling circuit30is disposed in the feeding line10AF and the parasitic radiating element10B is connected to the second coupling element32, since there is no correlation between the phase of the current iB flowing in the parasitic radiating element10B and the phase of the current iA flowing in the feeding radiating element10A, as described above, the resonance of the feeding radiating element10A and the resonance of the parasitic radiating element10B may have the same phase, in which case, the above notch is present. In the present preferred embodiment, such a problem does not arise, and the parasitic radiating element10B and the feeding radiating element10A are able to be provided adjacent to each other.

Although the present preferred embodiment is an example in which the feeding radiating element is a PIFA, the feeding radiating element is not limited to a PIFA and may be a typical inverse-F antenna. The same or substantially the same advantageous effects are able to be obtained.

Fifteenth Preferred Embodiment

A fifteenth preferred embodiment of the present invention will illustrate an example of an antenna device including a plurality of parasitic radiating elements.

FIG.31Aillustrates a circuit configuration of an antenna device115according to the fifteenth preferred embodiment. The antenna device115according to the present preferred embodiment includes the feeding radiating element10A, the feeding line10AF, the parasitic radiating element10B, a parasitic radiating element10C, and the coupling circuit30. The feeder circuit1is connected between the feeding line10AF and the ground.

The parasitic radiating element10C is, at around a ground end thereof, mainly coupled to the feeding line10AF of the feeding radiating element10A. The remaining configuration is the same as or similar to that of the antenna device114illustrated inFIG.30.

FIG.31Billustrates a frequency characteristic of a return loss of the antenna device115illustrated inFIG.31Aand an antenna device according to a comparative example. InFIG.31B, a return loss characteristic RL1is a return loss of the antenna device115according to the present preferred embodiment, and a return loss characteristic RL2is a return loss of the antenna device according to the comparative example. The antenna device according to the comparative example is an antenna device in which the coupling circuit30and the parasitic radiating element10B are not included and the first coupling element31merely defines and functions as a short pin of a PIFA. In either antenna device, a pole is generated at a center frequency F1of a low band. This is due to ¼ wavelength resonance of the feeding radiating element10A. Another pole is generated at a frequency F2. This is due to ¾ wavelength resonance of the feeding radiating element10A. A still another pole is generated at a frequency F3. This is due to ¼ wavelength resonance of the monopole parasitic radiating element10C.

In the antenna device115according to the present preferred embodiment, a pole is also generated at a frequency F0. This is due to a resonance characteristic of the parasitic radiating element10B. In this manner, it is possible to provide an antenna device including the parasitic radiating element10B that is connected to the coupling circuit30and the parasitic radiating element10C that does not interpose coupling of the coupling circuit30.

Also in the present preferred embodiment, the feeding radiating element is not limited to a PIFA and may be a typical inverse-F antenna. The same or substantially the same advantageous effects are obtained.

Sixteenth Preferred Embodiment

A sixteenth preferred embodiment of the present invention will illustrate an example of an antenna device including a plurality of parasitic radiating elements.

FIG.32illustrates a circuit configuration of an antenna device116according to the sixteenth preferred embodiment. The antenna device116according to the present preferred embodiment includes the feeding radiating element10A, the feeding line10AF, a short pin10AS, the parasitic radiating elements10B and10C, and the coupling circuit30. The feeding radiating element10A is a radiating element of a PIFA.

In the present preferred embodiment, the first coupling element31of the coupling circuit30is disposed around the ground end of the parasitic radiating element10B, and the second coupling element32of the coupling circuit30is disposed around the ground end of the parasitic radiating element10C. The parasitic radiating element10B is, at around the ground end thereof, mainly coupled to the feeding line10AF of the feeding radiating element10A.

As in the present preferred embodiment, the two parasitic radiating elements10B and10C may be configured to be coupled to each other via the coupling circuit30.

Note that in the present preferred embodiment, the feeding radiating element is not limited to a PIFA or an inverted-F antenna, and may be, for example, a monopole radiating element. That is, any feeding radiating element that is coupled to the parasitic radiating element10B may be used, and the same or substantially the same advantageous effects are obtained.

Seventeenth Preferred Embodiment

FIG.33illustrates a circuit configuration of an antenna device117according to a seventeenth preferred embodiment of the present invention. The antenna device117according to the present preferred embodiment includes feeding radiating elements10U and10V, the feeding line10AF, the parasitic radiating element10B, the parasitic radiating element10C, and the coupling circuit30. The feeder circuit1is connected between the feeding line10AF and the ground. The configuration and advantageous effects of the coupling circuit30are as described in the above preferred embodiments.

The feeding radiating elements10U and10V and the feeding line10AF define a branch-feeding monopole antenna or a branch-feeding PIFA. The parasitic radiating element10C is mainly coupled with the feeding line10AF to define and function as a monopole or an inverted-L antenna.

FIG.34illustrates a frequency characteristic of a return loss of the antenna device117. InFIG.34, a pole indicated by a frequency F1is mainly due to a fundamental wave generated in the feeding radiating element10U and the feeding line10AF in a branch antenna defined by the feeding radiating elements10U and10V and the feeding line10AF. A pole indicated by a frequency F2is due to a fundamental wave generated in the parasitic radiating element10C. A pole indicated by a frequency F3is mainly caused by, for example, a ¾ wavelength harmonic generated in the feeding radiating element10U and the feeding line10AF. A pole indicated by a frequency F4is due to a fundamental wave generated in the parasitic radiating element10B. A pole indicated by a frequency F5is mainly due to resonance generated in the feeding radiating element10V in the branch antenna defined by the feeding radiating elements10U and10V and the feeding line10AF.

Note that a parasitic capacitance is actively generated between the feeding radiating element10V and the parasitic radiating element10B so that a phase difference of the resonant current between the feeding radiating element10V and the parasitic radiating element10B is about 90°. Thus, a pole of the feeding radiating element10V indicated by the frequency F4and a pole of the parasitic radiating element10B indicated by the frequency F5are generated.

In the antenna device according to the present preferred embodiment, by including the branch antenna including the feeding radiating element10V, a communication band that is broadened to about 2700 MHz, for example, is able to be covered, and a broad-band antenna that covers a low band of about 700 MHz to about 900 MHz and a high band of about 1700 MHz to about 2700 MHz, for example, is able to be provided.

Eighteenth Preferred Embodiment

FIG.35is a circuit diagram of an antenna device118A according to an eighteenth preferred embodiment of the present invention. In the antenna device118A, the parasitic radiating element10B is provided at a side surface portion of the metal housing. The second coupling element32of the coupling circuit is connected to the parasitic radiating element10B. The remaining configuration is the same as or similar to that of the circuit illustrated inFIG.4in the first preferred embodiment.

With the structure of the antenna device118A, the parasitic radiating element10B is separated from the radiating element10, and a good radiation characteristic is able to be obtained at a resonant frequency that is added by the coupling circuit30and the parasitic radiating element10B. Furthermore, the radiation characteristic of the radiating element10is not degraded at frequencies other than the resonant frequency.

FIG.36is a circuit diagram of another antenna device118B according to the eighteenth preferred embodiment. In the antenna device118B, the parasitic radiating element10B is provided at a side surface portion of the metal housing. An end portion of the parasitic radiating element10B is connected to the ground (is grounded) of a circuit substrate, for example, via the inductor8. The parasitic radiating element10B defines and functions as a ½ wavelength resonant antenna.

With the structure of the antenna device118B, since the tip of the side surface portion of the metal housing is grounded, variations in antenna characteristic due to a change of surrounding environment is able to be reduced or prevented. Even in a case in which a side surface portion of another metal housing that is grounded via a slit is present forward of the tip of the side surface portion of the metal housing, since the tip of the side surface portion of the metal housing is grounded, a field maximum point moves from the tip of the parasitic radiating element10B toward a center, and a good radiation characteristic is able to be obtained at a resonant frequency that is added by the coupling circuit30. Furthermore, the resonant frequency is able to be easily adjusted by the inductance of the inductor8.

FIG.37is a circuit diagram of another antenna device118C according to the eighteenth preferred embodiment. In the antenna device118C, the feeding radiating element10A extends from an end surface portion of the metal housing to a side surface portion thereof. Similarly, the parasitic radiating element10B is extends from an end surface portion of the metal housing to a side surface portion thereof. In this manner, the main portion of the parasitic radiating element10B is provided at the end surface portion of the metal housing. In addition, the parasitic radiating element10B may be close to a ground end of the feeding radiating element10A. With this structure, since a field maximum point of the feeding radiating element10A moves from the ground end toward a center, unnecessary interference between the feeding radiating element10A and the parasitic radiating element10B is able to be reduced or prevented.

Nineteenth Preferred Embodiment

FIG.38is a plan view of a main portion of an antenna device119according to a nineteenth preferred embodiment of the present invention.

A metal housing of electronic equipment includes the radiating element10, defined by an end portion of the metal housing. A connection position of the feeding line9for the radiating element10and a position of the non-radiating resonant circuit20differ from those in the antenna device101illustrated inFIG.2in the first preferred embodiment.

In the present preferred embodiment, in a plan view of the circuit substrate6, the feeding line9is connected to the left side surface portion13of the radiating element10. Accordingly, the non-radiating resonant circuit20is disposed on the right side of the coupling circuit30. This positional relationship is an alternative configuration (symmetric relationship) to the example illustrated inFIG.2. The remaining configuration is the same as or similar to that illustrated in the first preferred embodiment.

FIG.39is a perspective view of the coupling circuit30according to the present preferred embodiment. The external configuration of the coupling circuit30is represented by a two-dotted-and-dashed line. On an outer surface of the coupling circuit30, the feeder circuit connection terminal PF, the radiating element connection terminal PA, the ground terminal PG, and the non-radiating resonant circuit connection terminal PS are formed. The coupling circuit30is the same or substantially the same as the coupling circuit30illustrated inFIG.1in the first preferred embodiment. However, the second surface MS2is the mount surface that faces the circuit substrate. On a top surface (first surface) that is opposed to the mount surface (second surface) MS2, the direction discrimination mark DDM is provided. Thus, the position of the terminals differ from that in the coupling circuit30illustrated inFIG.1in a plan view. In the coupling circuit30illustrated inFIG.1, in a plan view, the ground terminal PG, the feeder circuit connection terminal PF, and the non-radiating resonant circuit connection terminal PS are disposed clockwise in this order from the radiating element connection terminal PA. In the nineteenth preferred embodiment, as illustrated inFIG.39, the ground terminal PG, the feeder circuit connection terminal PF, and the non-radiating resonant circuit connection terminal PS are disposed counterclockwise in this order from the radiating element connection terminal PA.

As described above, since the first end and the second end of the first coupling element and the first end and the second end of the second coupling element are provided on both the first surface MS1and the second surface MS2, either the first surface or the second surface may define and function as the mount surface. Accordingly, either the first surface MS1or the second surface MS2of the coupling circuit30may be selected as the mount surface to be mounted on a circuit substrate such that the terminals are disposed at positions appropriate for the position of a circuit or an element to which the first coupling element and the second coupling element provided on the coupling circuit30are connected.

The examples illustrated inFIG.8andFIG.39illustrate examples in which interlayer connection conductors that connect the four terminals provided on the first surface MS1and the four terminals provided on the second surface MS2to each other are provided on end surfaces of the multi-layer body. However, a plurality of via conductors may be provided inside the multi-layer body, and the four terminals provided on the first surface MS1and the four terminals provided on the second surface MS2may be connected to each other via these via conductors.

In addition to formation of the above via conductors, LGA (Land Grid Array) terminals, for example, may preferably be provided on the mount surface of the coupling circuit30.

FIG.40illustrates a configuration of another coupling circuit30according to the present preferred embodiment and is an exploded plan view illustrating conductor patterns provided on layers of the coupling circuit30.

As illustrated inFIG.40, the first conductor pattern L11, the second conductor pattern L12, the third conductor pattern L21, and the fourth conductor pattern L22are respectively provided on the insulating material S11, the insulating material S12, the insulating material S21, and the insulating material S22. The insulating materials S11, S12, S21, and S22are stacked such that these coil conductor patterns are disposed in the following order from a layer close to the mount surface: the first conductor pattern L11, the second conductor pattern L12, the third conductor pattern L21, and the fourth conductor pattern L22.

A first end of the first conductor pattern L11is connected to the radiating element connection terminal PA, and a second end thereof is connected to a first end of the second conductor pattern L12via the interlayer connection conductor V1. A second end of the second conductor pattern L12is connected to the feeder circuit connection terminal PF. A first end of the third conductor pattern L21is connected to the non-radiating resonant circuit connection terminal PS, and a second end of the third conductor pattern L21is connected to a first end of the fourth conductor pattern L22via the interlayer connection conductor V2. A second end of the fourth conductor pattern L22is connected to the ground terminal PG.

The conductor patterns on the layers illustrated inFIG.40are preferably in a symmetric or substantially symmetrical relationship with the conductor patterns illustrated inFIG.9. Thus, in the coupling circuit including these conductor patterns, in a plan view, the ground terminal PG, the feeder circuit connection terminal PF, and the non-radiating resonant circuit connection terminal PS are disposed counterclockwise in this order from the radiating element connection terminal PA.

As in this example, the terminals may be disposed at positions appropriate for the position of a circuit or an element to which the first coupling element and the second coupling element provided in the coupling circuit30are connected.

Twentieth Preferred Embodiment

A twentieth preferred embodiment of the present invention will illustrate an antenna device further including a phase shifter.

FIG.41is a circuit diagram of an antenna device120according to the twentieth preferred embodiment in which the feeder circuit1is connected. In the antenna device120, a phase shifter50is connected between the feeder circuit1and the first coupling element31of the coupling circuit30. The phase shifter50is a phase shifter by which a phase shift amount changes depending on the frequency (has frequency dependency). The phase shifter50includes a first coil Lp, a second coil Ls, and a capacitor C3that are coupled to one another.

Note that in this example, capacitors C4and C5that provide impedance matching are connected between the feeder circuit1and the phase shifter50.

The configuration of the coupling circuit30, the radiating element10, and the non-radiating resonant circuit20is the same as or similar to that illustrated in the first preferred embodiment.

FIG.42is an equivalent circuit diagram illustrating the phase shifter50in which an ideal transformer IT and parasitic inductance components (series parasitic inductance components La and Lc and parallel parasitic inductance component Lb) are separately illustrated.

The coupling coefficient between the first coil Lp and the second coil Ls illustrated inFIG.41is lower than that of a common high-frequency transformer, and accordingly, the series parasitic inductance component Lc is large. However, since the capacitance of the capacitor C3is also large, impedance matching is ensured. In addition, since the capacitance of the capacitor C3is large, a ratio of a high-band signal bypassing the capacitor C3is higher than that bypassing the transformer defined by the first coil Lp and the second coil Ls, and a phase shifting effect of the transformer is small. On the other hand, for a low band, the amount bypassing the capacitor C3is relatively small, and the phase shifting effect of the transformer is large. Thus, the coupling coefficient is preferably determined such that the phase shift amount with respect to a low-band signal is about 180° and the phase shift amount with respect to a high-band signal is about 90°, for example.

FIG.43illustrates a frequency characteristic of the phase shift amount of the phase shifter50. In this example, for example, the phase shift amount in a low band (about 700 MHz to about 900 MHz band, for example) is preferably about 180°, and the phase shift amount in a high band (about 1.7 GHz to about 2.7 GHz band, for example) is preferably about 90°.

Next, effects obtained by providing the phase shifter50together with the coupling circuit30will be described.FIG.44Ais a circuit diagram of the antenna device illustrated in the first preferred embodiment, which does not include the phase shifter50, andFIG.44Billustrates impedance loci representing, on a Smith chart, impedances when seeing the antenna device from the feeder circuit1.

FIG.45Ais a circuit diagram of an antenna device to which the phase shifter50is included, andFIG.45Billustrates impedance loci representing, on a Smith chart, impedances when seeing the antenna device from the feeder circuit1. This antenna device is a circuit that does not include the capacitor C5in the circuit illustrated inFIG.41.

FIG.46Ais a circuit diagram of an antenna device including the impedance matching capacitor C5(the same as or similar to that illustrated inFIG.41), andFIG.46Billustrates an impedance locus representing, on a Smith chart, an impedance when seeing the antenna device from the feeder circuit1.

InFIG.44B, locus T0is an impedance locus of an antenna device according to a comparative example in which the coupling circuit30and the non-radiating resonant circuit20are not provided, and locus T1is an impedance locus of the antenna device illustrated inFIG.44A. Both are results obtained by a sweep from about 1.7 GHz to about 2.7 GHz. As is clear fromFIG.44B, by including the coupling circuit30and the non-radiating resonant circuit20, as described above, a pole (small loop on chart) is generated in the frequency characteristic of the antenna, and accordingly, the resonant frequency band moves toward the center of the chart. Note that a higher frequency band is present in a periphery of the chart, and it is discovered that matching is difficult in the high frequency band.

InFIG.45B, locus T2is an impedance locus of the antenna device including the phase shifter50, the coupling circuit30, and the non-radiating resonant circuit20, and locus T1is the same as or similar to locus T1illustrated inFIG.44A. Both are results obtained by a sweep from about 1.7 GHz to about 2.7 GHz. As is clear fromFIG.45B, by including the phase shifter50, the phase advances by about 180° in a low band, and the phase advances by about 90° in a high band. Accordingly, the high-frequency band also moves toward the center of the chart.

InFIG.46B, locus T3is an impedance locus of the antenna device illustrated inFIG.46A, and is a result obtained by a sweep from about 1.7 GHz to about 2.7 GHz. As is clear from comparison with locus T2illustrated inFIG.45B, by a function of the capacitor C5that is shunt-connected, the high-frequency band rotates clockwise. Thus, matching is improved in all frequency bands.

FIG.47illustrates a frequency characteristic of a return loss of the antenna devices illustrated inFIG.44AandFIG.46Aand the antenna device according to the comparative example. InFIG.47, a return loss characteristic RL1is a return loss characteristic of the antenna device according to the comparative example, in which the coupling circuit30and the non-radiating resonant circuit20are not included, a return loss characteristic RL2is a return loss characteristic of the antenna device illustrated inFIG.44A, and a return loss characteristic RL3is a return loss characteristic of the antenna device illustrated inFIG.46A. The return loss characteristics RL1and RL2inFIG.47are the same as those illustrated inFIG.6. Comparing the return loss characteristics RL2and RL3with each other, it is discovered that the return loss is small in all bands and that the high band is broadened to a wide band of, for example, from about 1.4 GHz to about 2.6 GHz, while using the same radiating element.

FIG.48is an external perspective view of the phase shifter50, andFIG.49is a plan view of layers in the phase shifter50. In addition,FIG.50is a sectional view of the phase shifter50.

A top surface of a material S1corresponds to a mount surface (bottom surface) of a multi-layer body100. On the material S1, a terminal T1as a first port P1, a terminal T2as a second port P2, a ground terminal G, and an open terminal NC are provided.

The material layers of the multi-layer body100may preferably be, for example, a non-magnetic ceramic multi-layer body made of LTCC or other suitable material or a resin multi-layer body made of a resin material, such as polyimide or liquid crystal polymer. In this manner, since the material layers are non-magnetic (not a magnetic ferrite), it is possible to use the material layers as a transformer and a phase shifter with a predetermined inductance and a predetermined coupling coefficient even in a high frequency band exceeding several hundreds of MHz.

Each of the conductor patterns and the interlayer connection conductors is preferably made of, for example, a conductor material including Ag or Cu as a main component and having a small resistivity. In a case in which the material layers are ceramic, for example, the conductor patterns and the interlayer connection conductors are preferably formed by screen printing and firing of a conductive paste including Ag or Cu as a main component. In a case in which the material layers are resin, for example, the conductor patterns and the interlayer connection conductors are preferably patterned by etching or other suitable method of a metal foil such as an Al foil or a Cu foil.

The phase shifter50includes a plurality of insulating materials S1to S9. Various conductor patterns are provided on the materials S1to S9. The “various conductor patterns” include not only conductor patterns provided on surfaces of the materials but also interlayer connection conductors. The interlayer connection conductors include not only via conductors but also end surface electrodes provided on end surfaces of the multi-layer body.

The top surface of the material S1corresponds to the mount surface (bottom surface) of the multi-layer body. On the material S1, the terminal T1as the first port P1, the terminal T2as the second port P2, the ground terminal G, and the open terminal NC are provided.

On the materials S5and S4, conductors L1A1and L1A2are provided, respectively. On the material S3, conductors L1A3and L1B1are formed. On the material S2, conductors L1B2and L1C are provided.

A first end of the conductor L1A1is connected to the terminal T1as the first port. A second end of the conductor L1A1is connected to a first end of the conductor L1A2via an interlayer connection conductor V11. A second end of the conductor L1A2is connected to a first end of the conductor L1A3via an interlayer connection conductor V12. A second end of the conductor L1A3is connected to a first end of the conductor L1B1. The second end of the conductor L1A3and the first end of the conductor L1B1are connected to a first end of the conductor L1B2via an interlayer connection conductor V13. A second end of the conductor L1B1is connected to a second end of the conductor L1B2via an interlayer connection conductor V14. The second end of the conductor L1B2is connected to a first end of the conductor L1C. A second end of the conductor L1C is connected to the ground terminal G.

On the materials S6and S7, conductors L2A1and L2A2are provided, respectively. On the material S8, conductors L2A3and L2B1are formed. On the material S9, conductors L2B2and L2C are provided.

A first end of the conductor L2A1is connected to the terminal T2as the second port. A second end of the conductor L2A1is connected to a first end of the conductor L2A2via an interlayer connection conductor V21. A second end of the conductor L2A2is connected to a first end of the conductor L2A3via an interlayer connection conductor V22. A second end of the conductor L2A3is connected to a first end of the conductor L2B1. The second end of the conductor L2A3and the first end of the conductor L2B1are connected to a first end of the conductor L2B2via an interlayer connection conductor V23. A second end of the conductor L2B1is connected to a second end of the conductor L2B2via an interlayer connection conductor V24. The second end of the conductor L2B2is connected to a first end of the conductor L2C. A second end of the conductor L2C is connected to the ground terminal G.

The conductors L1A1, L1A2, L1A3, L1B1, L1B2, and L1C and the interlayer connection conductors V11, V12, V13, and V14define the first coil Lp. In addition, the conductors L2A1, L2A2, L2A3, L2B1, L2B2, and L2C and the interlayer connection conductors V21, V22, V23, and V24define the second coil Ls. Both of the first coil Lp and the second coil Ls are preferably rectangular or substantially rectangular helical coils, for example.

Twenty-First Preferred Embodiment

A twenty-first preferred embodiment of the present invention will illustrate a radiating element having a structure that is different from that of the radiating element illustrated in the first preferred embodiment.

FIG.51is a plan view of a portion of a metal housing of electronic equipment. The metal housing of electronic equipment includes the radiating element10, which is defined by an end portion of the metal hosing, and the metal housing main portion40. Although the first preferred embodiment illustrated an example in which the end portion of the metal housing including three sides in a plan view is used as the radiating element10, as illustrated inFIG.51, the radiating element10may be defined by a planar end portion of the metal housing.

FIGS.52A and52Bare perspective views of portions of metal housings of different pieces of electronic equipment. In an example illustrated inFIG.52A, the radiating element10, which is defined by the end portion of the metal housing, includes a plane parallel or substantially parallel to the X-Y plane and a plane parallel or substantially parallel to the Y-Z plane. In an example illustrated inFIG.52B, the radiating element10, which is defined by the end portion of the metal housing, includes a plane parallel or substantially parallel to the X-Y plane, a plane parallel or substantially parallel to the Y-Z plane, and two planes parallel or substantially parallel to the X-Z plane.

As illustrated inFIGS.52A and52B, the end portion of the metal housing may have various shapes.

The above-described preferred embodiments have illustrated examples in which the end portion of the metal housing is used as the radiating element. However, a portion of the radiating element or the entire radiating element may be a conductor pattern provided on a circuit substrate, for example, or may be a member different from the housing.

Although the example illustrated inFIG.4illustrates a case in which a parasitic capacitance is generated between an end of the radiating element10and the ground, a capacitance having a low impedance in a high band may be actively provided at this position so as to cause the radiating element10to define and function as a PIFA. In addition, the position at which the capacitance is generated may be connected to the ground to define and function as a PIFA.

The linear conductor pattern of the non-radiating resonant circuit20is not limited to a shape that returns and may extend in one direction. Alternatively, the non-radiating resonant circuit20may be bent in an L-shape or may be curved, for example. Furthermore, the non-radiating resonant circuit20may include a conductor pattern that splits into a plurality of branches. Thus, a plurality of poles are able to be generated.

In addition, the non-radiating resonant circuit20may include a tip of the linear conductor pattern connected to the ground so as to define and function as a circuit similar to a short stub.

In the above-described examples, examples of using fundamental wave resonance of the non-radiating resonant circuit20have mainly described. However, any harmonic resonance of the non-radiating resonant circuit20, such as double-wave resonance (secondary resonance), triple-wave resonance (tertiary resonance), or 3/2-wave resonance, for example, may also be used. In addition, both of the fundamental wave resonance and the harmonic resonance may be used, or a plurality of harmonic resonances may be used.

As for the radiating element10, similarly, any harmonic resonance, such as double-wave resonance (secondary resonance), triple-wave resonance (tertiary resonance), or 3/2-wave resonance, for example, may also be used. In addition, both of the fundamental wave resonance and the harmonic resonance may be used, or a plurality of harmonic resonances may be used.

The above-described preferred embodiments have illustrated a smartphone or electronic equipment having the same shape as the smartphone. However, the preferred embodiments may be similarly applied to various types of electronic equipment, such as a mobile phone including a feature phone, a wearable terminal including a smart watch and smart glasses, a lap top PC, a tablet terminal, a camera, a game console, a toy, or other suitable devices, for example.

While preferred embodiments of the present invention have been described above, it is to be understood that variations and modifications will be apparent to those skilled in the art without departing from the scope and spirit of the present invention. The scope of the present invention, therefore, is to be determined solely by the following claims.