Patent ID: 12244255

DESCRIPTION OF EMBODIMENTS

Hereinafter, an electric motor drive device according to embodiments of the present disclosure and a refrigeration cycle application device including the electric motor drive device will be described with reference to the accompanying drawings.

First Embodiment

FIG.1is a diagram illustrating an exemplary configuration of a refrigeration cycle application device900according to a first embodiment. The refrigeration cycle application device900illustrated inFIG.1is an example of application of electric motor drive devices according to the first embodiment and second and third embodiments described later. The refrigeration cycle application device900includes an electric motor drive device200. Note that, althoughFIG.1illustrates an air conditioner of a separate type, the air conditioner is not limited to the separate type. Further, the refrigeration cycle application device900according to the first embodiment can be applied to a product including a refrigeration cycle, such as an air conditioner, a refrigerator, a freezer, or a heat pump water heater.

In the refrigeration cycle application device900illustrated inFIG.1, a compressor901, a four-way valve902, an indoor heat exchanger906, an expansion valve908, and an outdoor heat exchanger910are connected to each other via a refrigerant pipe912.

Inside the compressor901, a compression mechanism904that compresses a refrigerant, and an electric motor7that operates the compression mechanism904are provided. The electric motor drive device200is used to drive the electric motor7that is used for the compressor901.

The refrigeration cycle application device900can perform heating operation or cooling operation through switching operation of the four-way valve902. The compression mechanism904is driven by the electric motor7subjected to variable-speed control.

During the heating operation, as indicated by solid arrows, the refrigerant is pressurized and fed by the compression mechanism904, and returns to the compression mechanism904through the four-way valve902, the indoor heat exchanger906, the expansion valve908, the outdoor heat exchanger910, and the four-way valve902.

During the cooling operation, as indicated by broken arrows, the refrigerant is pressurized and fed by the compression mechanism904, and returns to the compression mechanism904through the four-way valve902, the outdoor heat exchanger910, the expansion valve908, the indoor heat exchanger906, and the four-way valve902.

During the heating operation, the indoor heat exchanger906acts as a condenser to release heat, and the outdoor heat exchanger910acts as an evaporator to absorb heat. During the cooling operation, the outdoor heat exchanger910acts as a condenser to release heat, and the indoor heat exchanger906acts as an evaporator to absorb heat. The expansion valve908decompresses and expands the refrigerant.

FIG.2is a schematic wiring diagram illustrating the electric motor drive device200according to the first embodiment, together with the electric motor7. The electric motor drive device200includes a booster circuit3, a smoothing capacitor5, a first voltage detector6, a power-supply current detector4, a second voltage detector10, a first control device8, a second control device100, an inverter80, a connection switching device60, a power supply circuit50as a control power supply, and a bus current detection unit40.

The booster circuit3includes a reactor2, a first leg31, and a second leg32. The first leg31and the second leg32are connected in parallel to each other. In the first leg31, a first upper-arm switching element311and a first lower-arm switching element312are connected in series. In the second leg32, a second upper-arm switching element321and a second lower-arm switching element322are connected in series. One end of the reactor2is connected to an alternating-current (AC) power supply1. Another end of the reactor2is connected to a connection point3abetween the first upper-arm switching element311and the first lower-arm switching element312in the first leg31. A connection point3bbetween the second upper-arm switching element321and the second lower-arm switching element322is connected to another end of the AC power supply1. In the booster circuit3, the connection points3aand3bconstitute an AC terminal. While converting an AC voltage output from the AC power supply1into a DC voltage, the booster circuit3boosts the DC voltage if necessary. Hereinafter, the voltage output from the AC power supply1is referred to as a “power supply voltage”. Note that the power supply voltage may be referred to as a “first voltage”.

Note that,FIG.2illustrates an example where the first and second upper-arm switching elements311and321and the first and second lower-arm switching elements312and322are metal oxide semiconductor field effect transistors (MOSFETs), but the switching elements are not limited to the MOSFETs. Instead of the MOSFETs, insulated gate bipolar transistors (IGBTs) may be used.

The first upper-arm switching element311includes a transistor and a diode connected in antiparallel to the transistor. Antiparallel means that a cathode of the diode is connected to a drain or a collector of the transistor, and an anode of the diode is connected to a source or an emitter of the transistor. When the first upper-arm switching element311is a MOSFET, a parasitic diode included in the MOSFET itself may be used as the diode. The parasitic diode is also referred to as a body diode. The other switching elements are similarly configured, and redundant description is omitted.

One end of the smoothing capacitor5is connected to a DC bus12aon a high potential side. The DC bus12ais drawn from a connection point3cbetween the first upper-arm switching element311in the first leg31and the second upper-arm switching element321in the second leg32. Another end of the smoothing capacitor5is connected to a DC bus12bon a low potential side. The DC bus12bis drawn from a connection point3dbetween the first lower-arm switching element312in the first leg31and the second lower-arm switching element322in the second leg32. In the booster circuit3, the connection points3cand3dconstitute a DC terminal.

As described above, the smoothing capacitor5is connected to the DC buses12aand12b. A boosted voltage output from the booster circuit3is applied across the smoothing capacitor5. The smoothing capacitor5smooths the output voltage of the booster circuit3. A bus voltage Vdc described above is a voltage smoothed by the smoothing capacitor5. That is, the booster circuit3boosts a voltage value of the bus voltage Vdc to be applied to the DC buses12aand12b.

The first voltage detector6is connected across the AC power supply1. The first voltage detector6detects a power supply voltage Vs. The power supply voltage Vs is an absolute value of an instantaneous voltage of the AC power supply1. A detected value of the power supply voltage Vs is input to the first control device8.

The power-supply current detector4is disposed between the AC power supply1and the booster circuit3. The power-supply current detector4detects a power supply current Is flowing between the AC power supply1and the booster circuit3. A detected value of the power supply current Is is input to the first control device8.

The second voltage detector10is connected across the smoothing capacitor5. The second voltage detector10detects the bus voltage Vdc. A detected value of the bus voltage Vdc is input to the first control device8.

The bus current detection unit40detects a bus current, that is, a direct current (DC) Idc flowing to an input side of the inverter80. The bus current detection unit40includes a shunt resistor inserted into the DC bus12b. A detected value of the bus current detection unit40is input to the second control device100.

The first control device8controls an operation of the booster circuit3. Specifically, the first control device8generates drive pulses for driving individual switching elements of the booster circuit3, on the basis of detected values of the first voltage detector6, the power-supply current detector4, and the second voltage detector10. Note that, when the drive pulses for driving individual switching elements of the booster circuit3are distinguished, a drive pulse for driving the first lower-arm switching element312may be referred to as a “first drive pulse”, while a drive pulse for driving the first upper-arm switching element311may be referred to as a “second drive pulse”. Furthermore, the drive pulses for driving the second upper-arm switching element321and the second lower-arm switching element322may be collectively referred to as “synchronous drive pulses”. The first drive pulse corresponds to Xa in the illustration, the second drive pulse corresponds to Xb in the illustration, and the synchronous drive pulses correspond to Ya and Yb in the illustration.

The second control device100controls operations of the inverter80and the connection switching device60. In order to control the inverter80, the second control device100generates pulse width modulation (PWM) signals Sm1to Sm6, and outputs the PWM signals Sm1to Sm6to the inverter80. In addition, in order to control the connection switching device60, the second control device100generates a connection selection signal Sc, and outputs the connection selection signal Sc to the connection switching device60.

The first control device8and the second control device100are implemented by a microprocessor. The microprocessor may be a processor or a processing device referred to as a central processing unit (CPU), a microcomputer, a digital signal processor (DSP), or the like. Further, there is no problem even if the first control device8and the second control device100are configured as one control device.

The electric motor7is an electric motor configured to enable switching a connection state of windings71,72, and73. An example of the electric motor7is a three-phase permanent magnet electric motor. Further, the connection switching device60includes switching units61,62, and63. In the electric motor7, end portions of the windings71,72, and73are drawn outside, and switching can be performed to either star connection (written as “Y connection” where appropriate) or delta connection (written as “A connection” where appropriate). This switching is performed by the switching units61,62, and63of the connection switching device60. That is, the connection switching device60performs an operation of mutually switching the connection state of the windings71,72, and73of the electric motor7between the Y connection and the A connection.

The power supply circuit50receives the bus voltage Vdc. The power supply circuit50steps down the received bus voltage Vdc to generate a control power supply voltage V8, a control power supply voltage V100, and a switching power supply voltage V60. The control power supply voltage V8is applied to the first control device8. The control power supply voltage V100is applied to the second control device100. The switching power supply voltage V60is applied to the connection switching device60.

Next, a basic circuit operation of the electric motor drive device200according to the first embodiment will be described with reference to the drawings ofFIGS.3to6.

FIG.3is a diagram illustrating a charging path for the smoothing capacitor5when the power supply voltage Vs has a positive polarity, in the booster circuit3ofFIG.2.FIG.4is a diagram illustrating a charging path for the smoothing capacitor5when the power supply voltage Vs has a negative polarity, in the booster circuit3ofFIG.2.FIG.5is a diagram illustrating a short-circuit path of the power supply voltage Vs through the reactor2when the power supply voltage Vs has a positive polarity, in the booster circuit3ofFIG.2.FIG.6is a diagram illustrating a short-circuit path of the power supply voltage Vs through the reactor2when the power supply voltage Vs has a negative polarity, in the booster circuit3ofFIG.2. Note that, the polarity of the power supply voltage Vs is defined as positive when a terminal on an upper side of the AC power supply1has a positive potential as illustrated inFIGS.3and5, and the polarity of the power supply voltage Vs is defined as negative when the terminal on the upper side of the AC power supply1has a negative potential as illustrated inFIGS.4and6.

When the switching operation of the first upper-arm switching element311, the first lower-arm switching element312, the second upper-arm switching element321, and the second lower-arm switching element322is not performed, as illustrated inFIG.3or4, a current for charging the smoothing capacitor5flows in accordance with the polarity of the power supply voltage Vs.

On the other hand, when the first lower-arm switching element312is turned ON while the power supply voltage Vs has a positive polarity, as illustrated inFIG.5, a short-circuit path can be formed in a path through the AC power supply1, the reactor2, the first lower-arm switching element312, the second lower-arm switching element322, and the AC power supply1. Further, when the first upper-arm switching element311is turned ON while the power supply voltage Vs has a negative polarity, as illustrated inFIG.6, a short-circuit path can be formed in a path through the AC power supply1, the second upper-arm switching element321, the first upper-arm switching element311, the reactor2, and the AC power supply1.

In the electric motor drive device200according to the first embodiment, switching control of these operation modes is performed under the control of the first control device8. The power supply current Is and the bus voltage Vdc can be switched by the switching control of the operation modes.

Note that the configuration of the booster circuit3illustrated inFIG.2is an example, and a booster circuit having a configuration other than that ofFIG.2may be used as long as the booster circuit has a booster function.

FIG.7is a block diagram illustrating an exemplary configuration of the first control device8according to the first embodiment. As illustrated inFIG.7, the first control device8includes a power supply current command value control unit20, a power supply current command value calculation unit21, an on-duty control unit22, a power supply voltage phase calculation unit23, an on-duty calculation unit24, a first drive pulse generation unit25, a second drive pulse generation unit26, and a synchronous drive pulse generation unit27.

The power supply current command value control unit20calculates a power supply current effective value command value Is_rms* on the basis of a deviation between the bus voltage Vdc detected by the second voltage detector10and a preset bus voltage command value Vdc*. The calculation of the power supply current effective value command value Is_rms* is achieved by performing proportional integral (PI) control on the deviation between the bus voltage Vdc and the bus voltage command value Vdc*. Note that the PI control is an example, and proportional (P) control or proportional integral differential (PID) control may be adopted instead of the PI control.

The power supply voltage phase calculation unit23estimates a power supply voltage phase estimated value es on the basis of the power supply voltage Vs detected by the first voltage detector6. The power supply voltage phase calculation unit23generates a sine value sin θs of the power supply voltage phase estimated value θs on the basis of the power supply voltage phase estimated value es.

The power supply current command value calculation unit21calculates a power supply current instantaneous value command value Is*. As in the illustration, the power supply current instantaneous value command value Is' can be obtained by multiplying the power supply current effective value command value Is_rms* output from the power supply current command value control unit20by the sine value sin θs of the power supply voltage phase estimated value θs output from the power supply voltage phase calculation unit23.

The on-duty control unit22calculates an on-duty DTa on the basis of the power supply current instantaneous value command value Is' and the power supply current Is. The on-duty DTa is a calculated duty value that is referred to when the first drive pulse Xa for turning ON the first lower-arm switching element312is generated.

The calculation of the on-duty DTa is performed by performing PI control on a deviation between the power supply current effective value command value Is_rms* and the power supply current Is. Note that the on-duty control unit22may also adopt P control or PID control instead of the PI control.

The on-duty calculation unit24calculates an on-duty DTb on the basis of the power supply voltage Vs, the bus voltage Vdc, and the on-duty DTa. The on-duty DTb is a calculated duty value that is referred to when the second drive pulse Xb for turning ON the first upper-arm switching element311is generated.

The first drive pulse generation unit25generates the first drive pulse Xa by comparing the on-duty DTa with an amplitude of a first triangular wave25awhich is a carrier wave. The second drive pulse generation unit26generates the second drive pulse Xb by comparing the on-duty DTb with an amplitude of a second triangular wave26awhich is a carrier wave. Note that the first triangular wave25aused in the first drive pulse generation unit25and the second triangular wave26aused in the second drive pulse generation unit26are 180° out of phase with each other.

The synchronous drive pulse generation unit27generates the synchronous drive pulse Ya by comparing the power supply voltage Vs with an amplitude of a third triangular wave27awhich is a carrier wave. Further, the synchronous drive pulse generation unit27generates the synchronous drive pulse Yb by comparing the power supply voltage Vs with an amplitude of a fourth triangular wave27bwhich is a carrier wave. The third triangular wave27aused to generate the synchronous drive pulse Ya and the fourth triangular wave27bused to generate the synchronous drive pulse Yb are 180° out of phase with each other.

FIG.8is a diagram illustrating an example of an operation of the power supply voltage phase calculation unit23illustrated inFIG.7.FIG.8illustrates waveforms of the power supply voltage Vs, the power supply voltage phase estimated value es, and the sine value sines of the power supply voltage phase estimated value es in order from the upper side. Note thatFIG.8illustrates waveforms under an ideal condition in which a delay due to control or a delay due to detection processing is not considered.

As illustrated inFIG.8, the power supply voltage phase estimated value θs is 360° at a point where the power supply voltage Vs is switched from the negative polarity to the positive polarity. The power supply voltage phase calculation unit23detects a point at which the power supply voltage Vs switches from the negative polarity to the positive polarity, and resets the power supply voltage phase estimated value θs, that is, returns the power supply voltage phase estimated value θs to 0°, at this switching point. Note that, in a case of using an interrupt function of a processor, a circuit that detects zero crossing of the power supply voltage Vs may be added toFIG.7. In any case, any method may be used as long as a phase of the power supply voltage Vs can be detected.

FIG.9is a diagram illustrating an exemplary configuration of the inverter80illustrated inFIG.2. As illustrated inFIG.9, the inverter80includes an inverter main circuit810and a drive circuit850, and an input terminal of the inverter main circuit810is connected to the DC buses12aand12b. The bus voltage Vdc is applied to the inverter main circuit810.

The inverter main circuit810includes six arm switching elements811to816. Freewheeling rectifying elements821to826are connected in antiparallel with the switching elements811to816.

The drive circuit850generates drive signals Sr1to Sr6on the basis of the PWM signals Sm1to Sm6. The drive signals Sr1to Sr6control ON/OFF of the switching elements811to816. At this time, an AC voltage having a variable frequency and a variable voltage value is generated and applied to the electric motor7. That is, the inverter80drives the electric motor7by applying the AC voltage having a variable frequency and a variable voltage value to the electric motor7, on the basis of the PWM signals Sm1to Sm6output from the second control device100.

The PWM signals Sm1to Sm6have magnitude (0 to 5 V) of a signal level of a logic circuit, whereas the drive signals Sr1to Sr6are signals having a voltage level necessary for controlling the switching elements811to816, for example, magnitude of +15 V to −15 V. Further, the PWM signals Sm1to Sm6use a ground potential of the second control device100as a reference potential, whereas the drive signals Sr1to Sr6each use a potential of an emitter terminal which is a negative terminal of a corresponding switching element as a reference potential.

Note that, althoughFIG.9illustrates a case where the switching elements811to816are IGBTs, the switching elements811to816are not limited thereto. Any element may be used as long as the element can perform the switching operation. Note that, in a case where the switching elements811to816are MOSFETs, since the switching elements811to816have a parasitic diode in structure, the switching elements811to816do not need to have the freewheeling rectifying element821to826.

As the switching elements811to816and the rectifying element821to826, a semiconductor element formed of a silicon-based material is generally used, but the switching elements811to816and the rectifying element821to826are not limited thereto. As the switching elements811to816and the rectifying element821to826, a switching element formed of a wide band gap (WBG) semiconductor of silicon carbide, gallium nitride, gallium oxide, diamond or the like may be used. By using the switching element and the rectifying element formed of a WBG semiconductor, a device with a lower loss can be configured.

FIG.10is a wiring diagram illustrating in detail a connection mode between the connection switching device60and the electric motor7illustrated inFIG.2.FIG.11is a diagram illustrating a detailed configuration of the switching units61,62, and63of the connection switching device60illustrated inFIG.2.

InFIG.10, first end portions71a,72a, and73aof the windings71,72, and73of three phases including a U-phase, a V-phase, and a W-phase of the electric motor7are connected to external terminals71c,72c, and73c, respectively. Further, the second end portions71b,72b, and73bof the windings71,72, and73of the U-phase, the V-phase, and the W-phase are connected to external terminals71d,72d, and73d, respectively. The external terminals71c,72c,73c,71d,72d, and73dare terminals that can be connected to a device external to the electric motor7. Output lines831,832, and833of the U-phase, the V-phase, and the W-phase of the inverter80are connected to the external terminals71c,72c, and73c.

As described above, the connection switching device60includes the switching units61,62, and63. Currents flowing through the windings71,72, and73flow through the switching units61,62, and63, respectively. The switching units61,62, and63switch paths of the currents flowing through the windings71,72, and73, respectively. As the switching units61,62, and63, electromagnetic contactors whose contacts are electromagnetically opened and closed are used. Such electromagnetic contactors include those referred to as a relay, a contactor, and the like. The switching units61,62, and63are configured as illustrated inFIG.11, for example. InFIG.11, the contacts of the switching units61,62, and63are configured to be in different states of connection when a current flows through excitation coils611,621, and631and when a current does not flow.

InFIG.11, the excitation coils611,621, and631are connected so as to receive the switching power supply voltage V60via a semiconductor switch604. Opening/closing of the semiconductor switch604is controlled by the connection selection signal Sc output from the second control device100. For example, the semiconductor switch604is turned OFF when the connection selection signal Sc has a first value, and the semiconductor switch604is turned ON when the connection selection signal Sc has a second value. The first value is, for example, a logical value “Low”, and the second value is, for example, a logical value “High”. The relationship of those may be reversed. Note that, in a case where the connection selection signal Sc is output from a circuit having a sufficient current capacity, a configuration may be adopted in which a current according to the connection selection signal Sc flows directly from the circuit to the excitation coils611,621, and631. In this case, the semiconductor switch604is unnecessary.

Note that the semiconductor switch604is generally formed using a semiconductor element formed of a silicon-based material, but is not limited thereto. As the semiconductor switch604, a semiconductor element formed of a WBG semiconductor may be used. By using a switching element formed of a WBG semiconductor, a device with a lower loss can be configured.

Returning toFIG.10, a common contact61cof the switching unit61is connected to the external terminal71dvia a lead wire61e, a normally closed contact61bis connected to a neutral point node64, and a normally open contact61ais connected to the V-phase output line832of the inverter80. A common contact62cof the switching unit62is connected to the external terminal72dvia a lead wire62e, a normally closed contact62bis connected to the neutral point node64, and a normally open contact62ais connected to the W-phase output line833of the inverter80. A common contact63cof the switching unit63is connected to the external terminal73dvia a lead wire63e, a normally closed contact63bis connected to the neutral point node64, and a normally open contact63ais connected to the U-phase output line831of the inverter80.

InFIG.11, when no current flows through the excitation coils611,621, and631, the switching units61,62, and63are in a state switched to the normally closed contact side as in the illustration, that is, the common contacts61c,62c, and63care in a state connected to the normally closed contacts61b,62b, and63b. In this state, the electric motor7is in the Y connection state. When a current flows through the excitation coils611,621, and631, the switching units61,62, and63are in a state switched to the normally open contact side contrary to the illustration, that is, the common contacts61c,62c, and63care in a state connected to the normally open contacts61a,62a, and63a. In this state, the electric motor7is in the Δ connection state.

Here, with reference toFIG.12, an advantage of using an electric motor capable of switching to either the Y connection or the A connection as the electric motor7will be described.FIG.12is a diagram illustrating two connection states to be switched in the electric motor7illustrated inFIG.2.

InFIG.12, (a) illustrates a state of connection when three windings are set to the Y connection, and (b) illustrates a state of connection when three windings are set to the Δ connection. It is assumed that VYis a line voltage at the time of the Y connection, IYis an inflowing current, VΔis a line voltage at the time of the A connection, and IΔis an inflowing current, and voltages applied to the windings of the individual phases are equal to each other. At this time, the relationship of the following Formula (1) is established between the voltage VYand the voltage VΔ.
VΔ=√VY/√3  (1)

In addition, the relationship of the following Formula (2) is established between the current IYand the current IΔ.
IΔ=√3×IY(2)

When the voltage VYand the current IYat the time of the Y connection and the voltage VΔand the current IΔat the time of the A connection have the relationships of the above Formulas (1) and (2), electric power supplied to the electric motor7is equal for the Y connection and the A connection. That is, when the electric powers supplied to the electric motor7are equal to each other, a current necessary for driving is larger in the Δ connection, and conversely, a voltage necessary for driving is lower in the Δ connection.

Using the properties described above, it is conceivable to select a connection state in accordance with a load condition or the like. For example, it is conceivable to operate at a low speed in the Y connection at the time of a low load, and operate at a high speed in the A connection at the time of a high load. By doing in such a way, the efficiency at the time of low load can be improved, and high output at the time of high load can be achieved.

This property will be described in detail by taking, as an example, a case where the electric motor7drives a compressor of an air conditioner.

As the electric motor7for driving a compressor in an air conditioner, a synchronous electric motor using a permanent magnet for a rotor is widely used in order to meet a demand for energy saving. Further, in air conditioners in recent years, when a difference between a room temperature and a set temperature is large, the electric motor7is rotated at a high speed to quickly bring the room temperature close to the set temperature. On the other hand, when the room temperature is close to the set temperature, the electric motor7is rotated at a low speed to maintain the room temperature. In a case of such an operation pattern, a proportion occupied by the time for driving at a low speed increases.

In a case of using the synchronous electric motor, when a rotation speed increases, a counter electromotive force increases and a voltage value necessary for driving increases. As described above, the counter electromotive force is higher in the Y connection than in the Δconnection.

In order to reduce the counter electromotive force in a high-speed rotation region, it is conceivable to reduce a magnetic force of the permanent magnet or reduce the number of turns of stator windings. However, in this case, since a current for obtaining the same output torque increases, a current flowing through the electric motor7and the inverter80increases; therefore, the efficiency of the device decreases.

For this reason, it is conceivable to switch the connection state in accordance with the rotation speed. For example, in the high-speed rotation region requiring high-speed operation, the connection state is set to the Δconnection. By doing in this way, a voltage value necessary for driving can be 1/√3 times that in the Y connection. This eliminates the need to reduce the number of turns of the windings and also eliminates the need to use magnetic flux weakening control.

On the other hand, in a low-speed rotation region in which operation can be performed at a low speed, the connection state is set to the Y connection. By doing in this way, a current value necessary for driving can be 1/√3 times that in the Δconnection. In addition, since the high-speed operation is not performed in the Y connection state, the windings in the Y connection state can be designed to be suitable for the low-speed driving. This makes it possible to reduce a current value as compared with a case where the Y connection is used over the entire speed range. As a result, a loss of the inverter80can be reduced, and the efficiency of the device can be increased.

Note that, in the high-speed rotation region, it is also conceivable to generate a voltage necessary for driving, by driving the booster circuit3and boosting the bus voltage Vdc. However, the efficiency is increased by operating in the Δ connection in which the necessary voltage is low rather than by boosting the bus voltage Vdc.

As described above, it is significant to switch the connection state of the windings71,72, and73of the electric motor7in accordance with the load condition. The connection switching device60is provided to enable such switching.

FIG.13is a block diagram illustrating an exemplary configuration of the second control device100according to the first embodiment. InFIG.13, the second control device100includes an operation control unit102and an inverter control unit110.

The operation control unit102receives command information Qefrom outside, and generates a stop signal St, the connection selection signal Sc, and a frequency command value ωe* on the basis of the command information Qe. The stop signal St is a signal for stopping the operation of the inverter80. The connection selection signal Sc is a signal for selecting the connection state of the windings71,72, and73described above. When a command value of a rotation speed of the electric motor7is “ωm*” and the number of pole pairs of the electric motor is “Pm”, the frequency command value ωe* can be obtained by ωe*=ωm*×Pm.

When controlling an air conditioner as the refrigeration cycle application device900, the second control device100controls an operation of each unit of the air conditioner on the basis of the command information Qe. The command information Qeis, for example, a temperature detected by a temperature sensor (not illustrated), information indicating a set temperature indicted by a remote controller which is an operation unit (not illustrated), operation mode selection information, instruction information for operation start and operation end, and the like. The operation mode selection information includes selection information based on the connection selection signal Sc in the case of the electric motor7in the first embodiment, in addition to, for example, heating, cooling, dehumidification, and the like. The operation control unit102may be external to the second control device100. That is, the second control device100may be configured to acquire the frequency command value ωe* from outside.

Here, a description is given to a change in the frequency command value ωe* and a behavior of the air conditioner when the Δ connection and the Y connection are selected in this order by the connection selection signal Sc. First, the Δ connection is selected at the time of activation, and the frequency command value ωe* gradually increasing to a frequency corresponding to a first target rotation speed after activation is generated. When the frequency command value ωe* reaches the frequency corresponding to the first target rotation speed, this state is maintained until the room temperature approaches the set temperature. When the room temperature comes close to the set temperature, the electric motor7is temporarily stopped and switched to the Y connection. After switching to the Y connection, reactivation is performed, and the frequency command value ωe* gradually increasing to a frequency corresponding to a second target rotation speed lower than the first target rotation speed is generated. After the frequency command value ωe* reaches the frequency corresponding to the second target rotation speed, control for maintaining a state where the room temperature is close to the set temperature is performed.

Next, the inverter control unit110will be described. As illustrated inFIG.13, the inverter control unit110includes a current restoration unit111, a three-phase/two-phase conversion unit112, an excitation current command value generation unit113, a voltage command value calculation unit115, an electrical angle phase calculation unit116, a two-phase/three-phase conversion unit117, and a PWM signal generation unit118.

The current restoration unit111restores phase currents iu, iv, and iwflowing through the electric motor7, on the basis of the direct current Idc detected by the bus current detection unit40. The current restoration unit111samples the direct current Idcdetected by the bus current detection unit40, at a timing determined on the basis of the PWM signals Sm1to Sm6generated by the PWM signal generation unit118. This allows the phase currents iu, iv, and iwto be restored from the direct current Idc.

The three-phase/two-phase conversion unit112converts the phase currents iu, ivand iwrestored by the current restoration unit111into a γ-axis current iγwhich is an excitation current and a δ-axis current iδwhich is a torque current, that is, current values of γ-δ axes, by using an electrical angle phase θegenerated by the electrical angle phase calculation unit116.

On the basis of the δ-axis current is, the excitation current command value generation unit113obtains a γ-axis current command value iγ* for achieving the best efficiency for driving the electric motor7. The γ-axis current command value iγ* for achieving the best efficiency is obtained when a current phase βmin which an output torque Tmis equal to or larger than a specified value or to be maximum is obtained, that is, when the current phase βmin which the current value is equal to or smaller than a specified value or to be minimum is obtained. Note that, inFIG.13, the γ-axis current command value iγ* is obtained on the basis of the δ-axis current is which is a torque current component, but the γ-axis current command value iγ* may be obtained on the basis of the γ-axis current iγ, and a frequency command value ω*.

On the basis of the frequency command value ωe* acquired from the operation control unit102, the γ-axis current iγand the δ-axis current iδacquired from the three-phase/two-phase conversion unit112, and the γ-axis current command value iγ* acquired from the excitation current command value generation unit113, the voltage command value calculation unit115generates a γ-axis voltage command value Vγ* and a δ-axis voltage command value Vδ*. Further, the voltage command value calculation unit115estimates a frequency estimation value west on the basis of the γ-axis voltage command value Vγ*, the δ-axis voltage command value Vδ*, the γ-axis current iγ, and the δ-axis current iδ. A detailed operation of the voltage command value calculation unit115will be described later.

The electrical angle phase calculation unit116calculates the electrical angle phase θeby integrating the frequency estimation value ωestacquired from the voltage command value calculation unit115.

The two-phase/three-phase conversion unit117converts the γ-axis voltage command value Vγ* and the δ-axis voltage command value Vδ′ acquired from the voltage command value calculation unit115, that is, voltage command values of a two-phase coordinate system, into three-phase voltage command values Vu*, Vv*, and VW* which are output voltage command values of a three-phase coordinate system, by using the electrical angle phase θe acquired from the electrical angle phase calculation unit116.

The PWM signal generation unit118compares the three-phase voltage command values Vu*, Vv*, and VW* acquired from the two-phase/three-phase conversion unit117with the bus voltage Vdc detected by the second voltage detector10, to generate the PWM signals Sm1to Sm6. Note that the PWM signal generation unit118can stop the electric motor7by not outputting the PWM signals Sm1to Sm6.

When the above-described stop signal St is generated by the operation control unit102, the generated stop signal St is provided to the PWM signal generation unit118. Upon receiving the stop signal St, the PWM signal generation unit118stops outputting the PWM signals Sm1to Sm6. As a result, the switching elements811to816of the inverter main circuit810stop the switching operation.

Note that, in the example described above, a configuration is adopted in which the phase currents iu, iv, and iware restored from the direct current Idc on the input side of the inverter80. However, a configuration may be adopted in which a current detector is provided in the output lines831,832, and833of the inverter80, and the phase current is detected by the current detector. In this case, the current detected by the current detector may be simply used instead of the current restored by the current restoration unit111.

Next, a description is given to an operation of the electric motor drive device200when the connection switching device60is operated during operation of the electric motor7.

First, problems of related art, that is, an operation in an electric motor drive device not including the features of the present disclosure will be described.

During operation of the electric motor7, that is, in a state where a current flows through the switching units61,62, and63constituting the connection switching device60, when the current flowing through the excitation coils611,621, and631is manipulated, connection of the common contacts61c,62c, and63cis switched between the normally closed contacts61b,62b, and63band the normally open contacts61a,62a, and63a, respectively. On the other hand, power supply from the inverter80to the electric motor7continues during operation of the electric motor7. Therefore, when the switching occurs, if a rotation speed of the electric motor7has not reached zero yet, arc discharge occurs between the individual contacts of the switching units61,62, and63. When the arc discharge occurs, there is a possibility that a failure such as contact welding occurs in the switching units61,62, and63.

In order to avoid such a failure, there is a control method of stopping power supply from the inverter80to the electric motor7to set the rotation speed of the electric motor7to zero before the connection switching device60is operated. By using this control method, it is possible to switch the connection state without causing arc discharge to occur between the individual contacts of the switching units61,62, and63.

On the other hand, when the rotation speed of the electric motor7is set to zero, it may be difficult to restart the electric motor7. For example, in a case where a load of the electric motor7is the compressor901, the state where the rotation speed is zero is a state where the refrigerant is not stable. In a case of restarting from this state, torque necessary for restarting increases and thus a necessary current also increases; therefore, there is a possibility that the restart cannot be performed, in the worst case. Therefore, it is necessary to restart the electric motor7after a lapse of time required for the state of the refrigerant to be sufficiently stabilized, without immediately operating the electric motor7. By doing in such a way, the refrigerant can no longer be pressurized by the compressor901, and the room temperature may be increased or decreased due to a decrease in air conditioning capacity, and the room temperature may not be kept constant.

Therefore, in the electric motor drive device200according to the present disclosure, control is performed such that a current flowing through the electric motor7or the connection switching device60during operation of the electric motor7is zero. Hereinafter, this control method is referred to as “zero current control”. When the connection switching device60is operated using the zero current control, it is possible to prevent arc discharge that may occur between the individual contacts of the switching units61,62, and63. This makes it possible to switch the connection state of the electric motor7without setting the rotation speed of the electric motor7to zero, that is, without stopping the electric motor7. When the zero current control is used, it is not necessary to stop the electric motor7before and after the switching of the connection state, so that a standby time for stabilizing the refrigerant of the compressor901is unnecessary. Therefore, it is possible to prevent an increase or a decrease in room temperature due to a decrease in air conditioning capacity. Note that, in the zero current control, the current flowing through the electric motor7or the connection switching device60does not need to be completely zero, and a state where the current is regarded as zero is sufficient.

Next, with reference to the drawings ofFIGS.14to16, a description will be given of a configuration and an operation of the voltage command value calculation unit115that implements the above-described zero current control.FIG.14is a diagram illustrating an exemplary configuration of the voltage command value calculation unit115that implements the zero current control in the first embodiment.FIG.15is a graph used to describe matters to be attended to when the zero current control in the first embodiment is performed.FIG.15illustrates a relationship between a modulation factor to be applied to the inverter80and an inverter output voltage which is an output voltage of the inverter80.FIG.16is a diagram illustrating an example of a control sequence when the zero current control in the first embodiment is performed.FIG.16illustrates waveforms of a current of the connection switching device60, the connection selection signal Sc, the bus voltage command value Vdc*, the frequency command value ω*, and a boosting operation of the booster circuit3, in order from the upper stage.

As illustrated inFIG.14, the voltage command value calculation unit115includes subtractors1151,1157, and1158, a frequency controller1152, current controllers1154and1156, switching units1153and1155, and a frequency estimation unit1159.

InFIG.14, the frequency estimation unit1159estimates a frequency of the electric motor7on the basis of the γ-axis current iγ, the δ-axis current iδ, the γ-axis voltage command value Vγ*, and the δ-axis voltage command value Vδ*, and generates the frequency estimation value ωest.

The subtractor1151obtains a difference value of the frequency estimation value ωestgenerated by the frequency estimation unit1159, with respect to frequency command value ω*. The difference value is a value of “ω*-ωest”.

The frequency controller1152performs, for example, PI control calculation on the difference value obtained by the subtractor1151, to obtain a δ-axis current command value iδ* for reducing the difference value. By generating such a δ-axis current command value iδ*, control for matching the frequency estimation value ωestwith the frequency command value ω* is performed.

The switching unit1153selects either the γ-axis current command value iγ* or the value 0, and outputs the selected command value to the subtractor1157as a γ-axis current command value iγ**. Further, the switching unit1155selects either the δ-axis current command value iδ* or the value 0, and outputs the selected command value to the subtractor1158as a δ-axis current command value iδ**. That is, either the γ-axis current command value iγ* or the value 0 is output from the switching unit1153by switching, and either the δ-axis current command value iδ* or the value 0 is output from the switching unit1155by switching.

The subtractor1157obtains a difference value of the output of the switching unit1153with respect to the γ-axis current iγ, and outputs the difference value to the current controller1154. The subtractor1158obtains a difference value of the output of the switching unit1155with respect to the δ-axis current is, and outputs the difference value to the current controller1156.

The current controller1154performs, for example, PI control calculation on the difference value obtained by the subtractor1157, to obtain the γ-axis voltage command value Vγ* for reducing the difference value. The current controller1156performs, for example, PI control calculation on the difference value obtained by the subtractor1158, to obtain the δ-axis voltage command value Vδ* for reducing the difference value.

When the γ-axis current command value iγ* is selected as the γ-axis current command value iγ* in the switching unit1153, control is performed to match the γ-axis current iγwith the γ-axis current command value iγ*. On the other hand, when the value 0 is selected as the γ-axis current command value iγ**, control is performed to set the γ-axis current iγto zero. Further, when the δ-axis current command value iδ* is selected as the δ-axis current command value iδ** in the switching unit1155, control is performed to match the 5-axis current iδwith the δ-axis current command value iδ*. On the other hand, when the value 0 is selected as the δ-axis current command value iδ**, control is performed to set the δ-axis current is to zero.

As illustrated inFIG.14, operating the switching unit1153to select the value 0 as the γ-axis current command value Iγ** and operating the switching unit1155to select the value 0 as the δ-axis current command value Iδ** are an example of a method for implementing the zero current control in the first embodiment.

Note that, as another example of the zero current control in the first embodiment, a method of stopping output of the PWM signals Sm1to Sm6is also considered. When output of the inverter80is stopped in a state where a large current flows through the windings71,72, and73of the electric motor7, there is a concern about generation of regenerative power or generation of a surge voltage. Therefore, it is also conceivable to switch a connection state of the windings71,72, and73by turning OFF all the outputs of the PWM signals Sm1to Sm6after performing the zero current control described above. In turning OFF the PWM signals Sm1to Sm6, similarly to the zero current control illustrated inFIG.14, no current flows when the following Formula (7) described below is satisfied. Note that, although other methods are conceivable for the zero current control, the description thereof will be omitted here.

Next, zero current control in consideration of an operation of the electric motor7will be described. First, a relationship between torque and speed in a general electric motor is expressed by the following Formula (3).
Δω=(Tm−Tl)/(Jm·(1/t))  (3)

In the above Formula (3), “Δω” represents a speed deviation, “Tm” represents an output torque, “Tl” represents a load torque, “Jm” represents an inertia moment, and “t” represents time.

Switching of the connection state of the windings71,72, and73requires a certain amount of time. Therefore, when switching the connection state of the windings71,72, and73, it is necessary to continue the zero current control for a certain period of time. The output torque is zero during the period of the zero current control. For this reason, as the period of the zero current control is longer and the load torque is larger, a reduction range of the speed becomes larger. Therefore, when the zero current control is started at a low speed, the rotation speed may drop to near zero, and the electric motor7may be out of step.

Therefore, it is conceivable to perform the zero current control after increasing the rotation speed of the electric motor7. However, as the rotation speed of the electric motor7increases, a counter electromotive force of the electric motor7increases, and thus it is necessary to output a voltage equal to or higher than the counter electromotive force from the inverter80. However, as illustrated inFIG.15, when the modulation factor to be applied to the inverter80exceeds one, a state enters a region where the inverter output voltage is saturated. A region where the modulation factor exceeds one is called an “overmodulation region”, and a region where the modulation factor is one or less is called a “non-overmodulation region”.

In the voltage saturation region where the inverter output voltage is saturated, it is necessary to perform magnetic flux weakening control in which a negative d-axis current flows, in order to reduce an apparent counter electromotive force of the electric motor7. However, when the zero current control is performed in this voltage saturation region, the negative d-axis current can no longer flow, and the counter electromotive force of the electric motor7becomes larger than a maximum output voltage of the inverter80to cause step out. In addition, in a case where the zero current control for turning OFF the PWM signal is performed, regeneration occurs, and the bus voltage Vdc becomes excessively large.

Therefore, when operating the connection switching device60to switch the connection state, the second control device100performs the zero current control after stopping a boosting operation of the booster circuit3. A more specific operation will be described with reference toFIG.16.

First, the bus voltage Vdc is boosted by the booster circuit3before the connection state is switched. When the boosting of the bus voltage Vdc is completed, the frequency command value ω* is increased. Then, immediately before switching the connection state, the boosting operation of the booster circuit3is stopped, and then the zero current control is performed. The connection state is switched during the zero current control, and the frequency command value ω* is lowered when the switching is completed. Note that, after connection switching control is finished, the boosting operation by the booster circuit3may be performed as necessary.FIG.16illustrates an example in which the boosting operation is performed after the connection switching control is finished. Note that,FIG.16is an example, and control may be performed in a sequence other thanFIG.16. However, in a case of performing the zero current control at a higher speed, it is necessary to increase the bus voltage Vdc and drive the electric motor7in the non-voltage saturation region.

The control described above will be supplemented. In order to stop the boosting operation of the booster circuit3, for example, it is sufficient if outputs of the first and second drive pulses Xa and Xb and the synchronous drive pulses Ya and Yb for controlling the operation of the booster circuit3are stopped. Note that this method is an example, and any control method may be used as long as the boosting operation of the booster circuit3can be stopped. In addition, the frequency command value ω* is determined such that the electric motor7rotates at a rotation speed of a degree that does not cause step out. Further, an output voltage of the booster circuit3, that is, the bus voltage Vdc is determined in accordance with the frequency command value ω*.

The above control enables non-stop connection switching control that is highly reliable.

Next, a description will be given of a set value of the bus voltage command value Vdc* at the time of performing the zero current control described above, by taking, as an example, a case where the electric motor7is a permanent magnet electric motor.

First, a voltage equation of a dq coordinate axis of the permanent magnet electric motor is expressed by the following Formulas (4) and (5).
Vd=(Ra+Ld·p)id−ωLqiq(4)
Vq=(Ra+Lq·p)iq+ωLdid+ωφa(5)

In the above Formulas (4) and (5), “Vd” and “Vq” represent dq-axis components of an armature voltage, “id” and “iq” represent dq-axis components of an armature current, “Ld” and “Lq” represent an inductance of a dq axis, “Ra” represents an armature winding resistance, “φa” represents an armature interlinkage magnetic flux of a permanent magnet in a dq coordinate system, and “p” represents a differential operator.

Further, in the above Formula (5), when id=iq=0 is established by the above-described zero current control, the following Formula (6) is obtained.
Vq=ωφa(6)

Therefore, in the case of performing the above-described zero current control, when the rotation speed of the electric motor7at the time of the connection switching is ω, the bus voltage Vdc needs to satisfy the following Formula (7).
Vdc≥√2·ω·φa(7)

Further, a value of the armature interlinkage magnetic flux φa changes depending on the connection state. For this reason, it is necessary to satisfy the above Formula (7) both before and after the connection switching. Therefore, the armature interlinkage magnetic flux φa needs to be set to a constant having a larger value of the armature interlinkage magnetic flux (pa before and after the connection switching.

For example, in a case of switching from the Δconnection to the Y connection, the armature interlinkage magnetic flux φa at the time of the Y connection is √3 times larger than that in the Δconnection, and thus a parameter of the Y connection is used for the armature interlinkage magnetic flux φa. Further, for example, in a case of switching from the Δ connection to the Y connection, when the bus voltage Vdc according to the above Formula (7) is set with the parameter at the time of the Δ connection as the armature interlinkage magnetic flux pa, a case is assumed in which the state enters the voltage saturation region at the time of the Y connection and the zero current control cannot be performed. In this case, since there is a possibility that an induced voltage of the electric motor7exceeds the bus voltage Vdc, the bus voltage Vdc is boosted so as to satisfy the above Formula (7).

With the above operation, it is possible to prevent a rapid increase in the bus voltage Vdc that may occur at the time of the connection switching. Further, during the connection switching, since the connection switching can be performed in a state where no current flows through the electric motor7and the switching units61,62, and63, it is possible to prevent occurrence of arc discharge between the individual contacts of the switching units61,62, and63. As a result, when a mechanical relay is used, contact welding can be prevented, and the electric motor drive device200with high reliability can be achieved. In addition, during the zero current control, since a current flowing through the switching units61,62, and63is set to zero, the connection switching can be performed without causing a large current change. This makes it possible to prevent a sudden change in rotation speed of the electric motor7due to the connection switching, and to perform the connection switching while reducing noise and vibration.

As described above, the electric motor drive device according to the first embodiment includes: the booster circuit; the inverter; the connection switching device that switches a connection state of windings of the electric motor; and the control device that controls operations of the booster circuit, the inverter, and the connection switching device. The control device performs zero current control to control the inverter such that a current flowing through the electric motor or the connection switching device is zero. Further, when operating the connection switching device to switch the connection state, the control device performs the zero current control after stopping a boosting operation of the booster circuit. By these controls, it is possible to prevent a rapid increase in bus voltage that may occur at a time of connection switching, while preventing occurrence of arc discharge between the individual contacts of the individual switching units. This can prevent damage to the switching element. In addition, since stress accumulation on the switching element can be prevented, it is possible to extend the life of the device and improve the reliability of the device. Further, to put it differently from another viewpoint, the connection switching device can be configured with a less expensive component in order to achieve desired durability, so that an increase in manufacturing cost of the device can be prevented.

In addition, in the electric motor drive device according to the first embodiment, in a case where the electric motor is a permanent magnet electric motor, the relationship of the above Formula (7) is desirably satisfied when the connection state of the windings is switched. When the relationship of the above Formula (7) is satisfied, a reduction range of the speed when the zero current control is performed can be reduced. As a result, even when the period of the zero current control becomes long, step out of the electric motor can be prevented.

Further, in the electric motor drive device according to the first embodiment, as pa used in the above Formula (7), it is desirable to use either the armature interlinkage magnetic flux φa before switching of the connection state or the armature interlinkage magnetic flux pa after of switching the connection state, whichever has a larger value. By using the armature interlinkage magnetic flux having a larger value, it is possible to prevent any one of the connection states before and after the connection state switching from entering the voltage saturation region. This allows the zero current control to be reliably performed.

Further, in the electric motor drive device according to the first embodiment, the control device desirably controls an output voltage of the booster circuit in accordance with a rotation speed of the electric motor at the time of switching the connection state. This allows the zero current control to be efficiently performed.

Second Embodiment

FIG.17is a wiring diagram illustrating in detail a connection mode between the connection switching device60and the electric motor7in the second embodiment. In the configuration ofFIG.10, a changeover switch is used as each of the switching units61,62, and63of the connection switching device60. Instead of that configuration, each switching unit may be configured by a combination of a normally closed switch and a normally open switch, and an exemplary configuration thereof is illustrated inFIG.17.

In the configuration ofFIG.17, instead of the changeover switch of the switching unit61, a combination of a normally closed switch615and a normally open switch616is used. Further, instead of the changeover switch of the switching unit62, a combination of a normally closed switch625and a normally open switch626is used. Further, instead of the changeover switch of the switching unit63, a combination of a normally closed switch635and a normally open switch636is used.

As in the illustration, in a state where the normally closed switches615,625, and635are closed and the normally open switches616,626, and636are opened, the electric motor7is in Y connection. Contrary to the illustration, in a state where the normally closed switches615,625, and635are opened and the normally open switches616,626, and636are closed, the electric motor7is in Δconnection.

As illustrated inFIG.17, even in a case where each switching unit of the connection switching device60is configured by a combination of a normally closed switch and a normally open switch, an electromagnetic contactor can be used as each switch. The electromagnetic contactor is suitable because a conduction loss at the time of ON is small.

Further, as the normally closed switches615,625, and635and the normally open switches616,626, and636illustrated inFIG.17, a switching element formed of a WBG semiconductor may be used. A semiconductor switch formed of a WBG semiconductor may be used. Since the semiconductor switch formed of a WBG semiconductor has a small ON-resistance, an effect of low loss and less element heat generation can be obtained. By using the semiconductor switch formed of a WBG semiconductor, the switching operation can be quickly performed.

As described above, the switching operation can be performed at a high speed even when the semiconductor switch is used, but operation variation of about several μs occurs between the individual semiconductor switches. Therefore, when a time constant L/R based on a winding resistance R and a winding inductance L of the electric motor7is very small, there is a possibility that a rapid current change occurs and the rotation speed of the electric motor7rapidly changes. As a result, there is a concern that vibration or noise is generated in the electric motor7, and the semiconductor switch generates heat to cause thermal destruction.

On the other hand, by applying the zero current control described in the first embodiment to the connection switching device60configured by the semiconductor switch, and performing the connection switching during execution of the zero current control, it is possible to perform connection switching without causing a large current change. This makes it possible to prevent a sudden change in rotation speed of the electric motor7at the time of the connection switching, and thus it is possible to switch the connection state while reducing noise or vibration. In addition, since it is possible to prevent a rapid change in rotation speed of the electric motor7at the time of the connection switching, thermal destruction caused by heat generation of the semiconductor switch can be prevented.

Third Embodiment

In the first and second embodiments, the zero current control in the first embodiment is applied to the electric motor7in which the connection state of the windings can be switched mutually between the Y connection and the Δconnection, but the electric motor7may be an electric motor in which a connection state of windings is another different connection mode. For example, an electric motor used may include two or more winding portions as a winding of each phase and be switchable to either parallel connection or series connection. In this case, it suffices that both end portions of each of the two or more winding portions constituting the winding of each phase are drawn outside the electric motor, and connection of the drawn both end portions of each of the two or more winding portions is switched. Hereinafter, a specific exemplary configuration and an operation in the exemplary configuration will be described.

FIG.18is a wiring diagram illustrating in detail a connection mode between the connection switching device60and an electric motor7A in the third embodiment.FIG.18illustrates a configuration in which, in the electric motor7A in Y connection, a winding of each phase is configured by two winding portions, both end portions of each of the winding portions are made connectable to the outside of the electric motor7, and a connection state is switched by the connection switching device60.

Specifically, the U-phase winding71includes two winding portions711and712, the V-phase winding72includes two winding portions721and722, and the W-phase winding73includes two winding portions731and732.

First end portions of the winding portions711,721, and73lare connected to the output lines831,832, and833of the inverter80via the external terminals71c,72c, and73c. Second end portions of the winding portions711,721, and73lare connected to common contacts of changeover switches617,627, and637via external terminals71g,72g, and73g.

First end portions of the winding portions712,722, and732are connected to common contacts of changeover switches618,628, and638via external terminals71h,72h, and73h. Second end portions of the winding portions712,722, and732are connected to the neutral point node64via the external terminals71d,72d, and73d.

Normally closed contacts of the changeover switches617,627, and637are connected to normally closed contacts of the changeover switches618,628, and638. Normally open contacts of the changeover switches617,627, and637are connected to the neutral point node64. Normally open contacts of the changeover switches618,628, and638are connected to the output lines831,832, and833of the inverter80. The changeover switches617,627,637,618,628, and638constitute the connection switching device60.

Even in a case where such a connection switching device60is used, the connection switching device60can be protected similarly to the first embodiment and the second embodiment.

In a case of the configuration illustrated inFIG.18, in a state where the changeover switches617,627,637,618,628, and638are switched to the normally closed contact side as in the illustration, the electric motor7A is in the series connection state. On the other hand, in a state where the changeover switches617,627,637,618,628, and638are switched to the normally open contact side contrary to the illustration, the electric motor7A is in the parallel connection state.

Note that, also in the third embodiment, as described in the second embodiment, a combination of a normally closed switch and a normally open switch can be used instead of the changeover switch.

Although a case has been described above in which switching between the series connection state and the parallel connection state is performed in the electric motor7A in the Y connection, the present disclosure is not limited to this example. The configuration according to the third embodiment is also applicable to switching between the series connection state and the parallel connection state, for example, in an electric motor in Δconnection.

In addition, the configuration has been described above in which the windings of individual phases of the Y connection or the Δ connection are switched to parallel connection or series connection, but the present disclosure is not limited to these examples. The configuration according to the third embodiment may be a configuration in which an intermediate tap is provided in the windings in the Y connection state or the Δ connection state, and a voltage necessary for driving is changed by short-circuiting some of the windings with a switching means. In short, the content of the present disclosure is applicable to any configuration as long as a connection state of windings of an electric motor can be switched, and a counter electromotive force can be switched by the switching of the connection state.

Note that the configuration described in the above embodiments is an example of the configuration of the present disclosure, and can be combined with another known technique, and it is obvious that a change such as partial omission can be made in the configuration without departing from the subject matter of the present disclosure.

REFERENCE SIGNS LIST

1AC power supply;2reactor;3booster circuit;3a,3b,3c,3dconnection point;4power-supply current detector;5smoothing capacitor;6first voltage detector;7,7A electric motor;8first control device;10second voltage detector;12a,12bDC bus;20power supply current command value control unit;21power supply current command value calculation unit;22on-duty control unit;23power supply voltage phase calculation unit;24on-duty calculation unit;25first drive pulse generation unit;25afirst triangular wave;26second drive pulse generation unit;26asecond triangular wave;27synchronous drive pulse generation unit;27athird triangular wave;27bfourth triangular wave;31first leg;32second leg;40bus current detection unit;50power supply circuit;60connection switching device;61,62,63switching unit;61e,62e,63elead wire;61a,62a,63anormally open contact;61b,62b,63bnormally closed contact;61c,62c,63ccommon contact;64neutral point node;71,72,73winding;71a,72a,73afirst end portion;71b,72b,73bsecond end portion;71c,71d,71g,71h,72c,72d,72g,72h,73c,73d,73g,73hexternal terminal;80inverter;100second control device;102operation control unit;110inverter control unit;111current restoration unit;112three-phase/two-phase conversion unit;113excitation current command value generation unit;115voltage command value calculation unit;116electrical angle phase calculation unit;117two-phase/three-phase conversion unit;118PWM signal generation unit;200electric motor drive device;311first upper-arm switching element;312first lower-arm switching element;321second upper-arm switching element;322second lower-arm switching element;604semiconductor switch;611,621,631excitation coil;615,625,635normally closed switch;616,626,636normally open switch;617,618,627,628,637,638changeover switch;711,712,721,722,731,732winding portion;810inverter main circuit;811to816switching element;821to826rectifying element;831,832,833output line;850drive circuit;900refrigeration cycle application device;901compressor;902four-way valve;904compression mechanism;906indoor heat exchanger;908expansion valve;910outdoor heat exchanger;912refrigerant pipe;1151,1157,1158subtractor;1152frequency controller;1153,1155switching unit;1154,1156current controller;1159frequency estimation unit.