Patent ID: 12216203

DESCRIPTION OF SPECIFIC EMBODIMENTS

The above known technique, as disclosed in JP 1988-249071 A, needs a large amount of calculation for correlation detection between the received and reference signals. Thus, an object detection device that uses such correlation detection is prone to have an increased circuit size.

In view of the foregoing, it is desired to have an object detection device capable of reducing the amount of calculation for correlation detection.

According to one aspect of the present embodiment, an object detection device includes a transceiver that transmits ultrasonic waves encoded with frequency modulation and receives an ultrasonic wave and outputs a reception signal, a first quadrature detector that generates and outputs a complex reception signal based on quadrature detection of the reception signal, a second quadrature detector that generates and outputs a complex reference signal based on quadrature detection of the reference signal, a correlation filter that performs correlation detection between the complex reception signal and the complex reference signal and outputs a correlation signal, and a code determiner that determines a code included in the reception signal based on the correlation signal.

In this manner, converting the reception signal and the reference signal into complex signals based on quadrature detection allows a correlation between them to be calculated using vector and matrix operations, thereby reducing the amount of calculation for correlation detection.

Hereinafter, some embodiments of the disclosure will be described with reference to the drawings. In order to facilitate understanding of the description, the same structural elements in the drawings share the same reference numerals, and repeated description is omitted.

First Embodiment

A first embodiment will now be described. An object detection device1of the present embodiment illustrated inFIG.1is mounted to a vehicle which is not shown in the figure, and is configured to detect an object B around the vehicle. The vehicle carrying the object detection device1is hereinafter referred to as an “own vehicle”. The vehicle (not shown) is, for example, an automobile.

The object detection device1includes an ultrasonic sensor2and a controller3that controls the operation of the ultrasonic sensor2. The ultrasonic sensor2is configured to detect an object B by transmitting probe waves which are ultrasonic waves, and receiving reflected waves of the probe waves by the object B.

The ultrasonic sensor2includes a transceiver4, a drive signal generator5, a matched filter6, a reference signal processor7, and a determiner8. The transceiver4includes a transmission section40A and a reception section40B. The transmission section40A is provided to enable transmission of probe waves to the outside. The reception section40B is provided to receive ultrasonic waves, including reflected waves by the object B, of the probe waves transmitted from the transmission section40A.

The transceiver4includes a transducer41, a transmission circuit42, and a reception circuit43. The transmitter section40A is formed of the transducer41and the transmission circuit42. The reception section40B is formed of the transducer41and the reception circuit43.

The transducer41serves as a transmitter to transmit the probe waves to the outside and as a receiver to receive the reflected waves, and is electrically connected to the transmission circuit42and the reception circuit43. That is, the ultrasonic sensor2has a so-called integrated transmitter/receiver configuration.

Specifically, the transducer41is configured as an ultrasonic microphone with a built-in electrical-mechanical energy conversion element, such as a piezoelectric element. The transducer41is disposed in a position facing the outer surface of the own vehicle so as to be capable of transmitting probe waves to the outside of the own vehicle and receiving reflected waves from the outside of the own vehicle.

The transmission circuit42is provided to drive the transducer41based on the drive signal received, thereby causing the transducer41to transmit a probe wave. Specifically, the transmission circuit42includes a digital-to-analog conversion circuit and the like. That is, the transmission circuit42is configured to generate an element input signal by performing signal processing such as digital-to-analog conversion on the drive signal output from the drive signal generator5. The element input signal is an AC voltage signal to drive the transducer41. The transmission circuit42is configured to apply the generated element input signal to the transducer41to excite the electrical-mechanical energy conversion element in the transducer41, thereby generating a probe wave.

The reception circuit43is configured to generate a reception signal corresponding to a result of reception of an ultrasonic wave by the transducer41and output the reception signal to the matched filter6. Specifically, the reception circuit43includes an amplification circuit and an analog-to-digital conversion circuit. That is, the reception circuit43is configured to perform signal processing, such as amplification and analog-to-digital conversion and the like, on an element output signal output from the transducer41to generate a reception signal that includes information about the amplitude and frequency of the received wave. The element output signal is an alternating voltage signal generated by the electrical-mechanical energy conversion element in the transducer41through reception of the ultrasonic wave.

As described later, the probe wave includes an ultrasonic wave encoded by frequency modulation. The center frequency of the frequency modulation band of the probe wave is fc, and the sampling frequency of the reception circuit43is at least twice fc. The sampling frequency of the reception signal may be the same as or different from the sampling frequency of the drive signal.

The drive signal generator5is configured to generate a drive signal and output it to the transmission circuit42. The drive signal is a signal for driving the transducer41to cause the transducer41to transmit a probe wave.

The drive signal generator5is configured to generate a drive signal corresponding to a frequency modulation state of the probe wave among predefined frequency modulation states. The drive signal generator5generates the drive signal such that the frequency of the probe wave is swept in a range including a resonant frequency of the transducer41.

In the present embodiment, the predefined frequency modulation states include an up-chirp or down-chirp. The up-chirp is a frequency modulation state such that the frequency increases monotonically with time. The down-chirp is a frequency modulation state such that the frequency decreases monotonically with time.

The drive signal generator5, the matched filter6, the reference signal processor7, and the determiner8may be configured, for example, as a Digital Signal Processor (DSP) having functions programmed, such as the above-described drive signal generation, as well as quadrature detection, correlation detection, code determination, object detection determination, and the like as described later.

The matched filter6processes the reception signal and performs correlation detection between the reception signal and a reference signal. The matched filter6includes a quadrature detector61and a correlation filter62. The quadrature detector61, which corresponds to a first quadrature detector, generates a complex signal based on quadrature detection of the reception signal output from the reception circuit43. As illustrated inFIG.2, the quadrature detector61includes a multiplier611, a low-pass filter (LPF)612, and a down-sampler613.

The multiplier611multiplies the reception signal output from the reception circuit43by sin(2πfct) and cos(2πfct) to generate a complex signal. Here, t is time. The signals sin(2πfct) and cos(2πfct) are input from the drive signal generator5to the multiplier611. The multiplier611outputs the generated complex signal to the LPF612.

The LPF612removes high frequency components from the complex signal output from the multiplier611. The cut-off frequency of the LPF612is received from the controller3and is set based on the bandwidth of the transducer41and the sweep band of the drive signals. The complex signal having the high-frequency components removed by the LPF612is input to the down-sampler613.

The down-sampler613down-samples the output signal from the LPF612. The down-sampler613, for example, down-samples the signal sampled at twice the center frequency fc to one times the center frequency fc. The sampling frequency after down-sampling may be set lower than one times the center frequency fc according to the cut-off frequency of the LPF612. Since the high frequency components have been removed by the LPF612, the reception signal can be down-sampled, which can reduce an amount of calculation by the correlation filter62, thereby reducing the circuit area.

In a configuration where the object detection device1includes a plurality of identical transceivers4and discriminates between a direct wave which occurs in the case of the transceiver4on the transmission side and the transceiver4on the reception side being the same, and an indirect wave which occurs in the case of the transceiver4on the transmission side and the transceiver4on the reception side being different, correlation calculations are required for both up- and down-chirps.

For example, in a case where one of two transceivers4transmits an up-chirp signal and the other transmits a down-chirp signal, a correlation calculation between the reception signal received at each of the two transceivers4and each of the two chirp signals is made. If the correlation between the reception signal at one of the transceiver units4that transmitted the up-chirp signal and the up-chirp signal is high, the ultrasonic wave received by this transceiver unit4is determined to be a direct wave. If the correlation with the down-chirp signal is high, this ultrasonic wave is determined to be an indirect wave. If the correlation between the reception signal at the other transceiver unit4that transmitted the down-chirp signal and the down-chirp signal is high, the ultrasonic wave received by this transceiver unit4is determined to be a direct wave. If the correlation with the up-chirp signal is high, this ultrasonic wave is determined to be an indirect wave.

Making multiple correlation calculations in this manner increases the amount of calculation, leading to an increased circuit size of the object detection device1. In contrast, down-sampling as described above can reduce the amount of calculation, leading to a reduced circuit size. The output signal of the down-sampler613is input to the correlation filter62.

The complex signal output from the down-sampler613is a complex reception signal. The complex reception signal consists of N signals sampled by the down-sampler613. N is an integer greater than or equal to 2. The N signals forming the complex reception signal are denoted as signals S1to SNin the order in which they were sampled.

The correlation filter62performs correlation detection between the complex reception signal generated by the quadrature detector61and each of the reference signals corresponding to the up-chirp and down-chirp, respectively, and outputs a correlation signal. The correlation signal output from the correlation filter62is input to the determiner8. Details of the correlation filter62will be described later.

The reference signal processor7processes signals output from the drive signal generator and outputs them to the matched filter6. The signals output from the drive signal generator5to the reference signal processor7correspond to the up-chirp and down-chirp used for the drive signal to be input to the transceiver4, where these signals are reference signals for identifying the code of the reception signal. The drive signal generator5outputs the reference signal corresponding to the up-chirp and the reference signal corresponding to the down-chirp to the reference signal processor7. In the matched filter6, the reference signals processed by the reference signal processor7are used for correlation detection. As illustrated inFIG.1, the reference signal processor7includes a quadrature detector71.

The quadrature detector71generates a complex signal based on quadrature detection of the reference signal output from the drive signal generator5, and corresponds to a second quadrature detector. As illustrated inFIG.3, the quadrature detector71includes a multiplier711, an LPF712, and a down-sampler713. The multiplier711, the LPF712, and the down-sampler713have the same configuration as the multiplier611, LPF612, and down-sampler613of the quadrature detector61.

That is, the multiplier711multiplies the reference signal by each of sin(2πfct) and cos(2πfct) to generate a complex signal, and the LPF712removes high-frequency components from the complex signal output from the multiplier711. The down-sampler713down-samples the output signal of the LPF712.

The down-sampler713performs down-sampling such that the sampling frequency after down-sampling for the reference signal is the same as the sampling frequency after down-sampling for the reception signal. That is, for example, if the input signal is down-sampled at one times the center frequency fc in the down-sampler613, the input signal is also down-sampled at one times the center frequency fc in the down-sampler713. Since this down-sampling performed is after the high-frequency components have been removed by LPF712, the reference signal can be down-sampled, which can reduce the amount of calculation for the correlation filter62and thus the circuit area can be reduced. The output signal of the down-sampler713is then input to the correlation filter62.

The complex signal output from the down-sampler713is a complex reference signal. The complex reference signal consists of N signals like the complex reception signal. The N signals forming the complex reference signal are the signals SR1to SRNin the order in which they are sampled. In the correlation filter62, correlation detection is performed between the complex reception signal, which consists of signals S1to SN, and the complex reference signal, which consists of signals SR1to SRN.

As illustrated inFIG.4, the correlation filter62includes an up-chirp filter620A and a down-chirp filter620B. The up-chirp filter620A performs correlation detection between the complex reception signal and the complex reference signal for the up-chirp signal. The down-chirp filter620B performs correlation detection between the complex reception signal and the complex reference signal for the down-chirp signal.

The up-chirp filter620A includes a reference signal holder621, a vector rotator622, an integrator623, and an amplitude converter624.

The up-chirp filter620A is configured to receive from the reference signal processor7the complex reference signal generated through quadrature detection of the reference signal corresponding to the up-chirp. The reference signal holder621is configured to hold and output the complex reference signal received from the reference signal processor7, and output the plurality of signals forming the complex reference signal individually. Specifically, the reference signal holder621outputs the signals SR1to SRNoutput from the down-sampler713individually. The vector rotator622performs vector rotation of the received signal. As illustrated inFIG.5, the vector rotator622includes a matrix converter625, a reception signal holder626, and a multiplier627.

The matrix converter625converts the signals SR1to SRNoutput from the reference signal holder621into rotation matrices R1to RN. Specifically, with the phase of signal SR1as θR1, the rotation matrix R1is generated as follows.

R1=[cos⁢θR⁢1sin⁢θR⁢1-sin⁢θR⁢1cos⁢θR⁢1]

The rotation matrices R2to RNare generated in the same way using the phases θR2to θRNof the signals SR2to SRN. The matrix converter625outputs the signals corresponding to the generated rotation matrices R1to RNindividually to the multiplication unit627.

The reception signal holder626holds the complex reception signal and outputs it to the multiplier627. The reception signal holder626is configured to receive the complex reception signal from the quadrature detector61, and the reception signal holder626outputs the received signals S1to SNindividually to the multiplier627.

The multiplier627multiplies the rotation matrices R1to RNgenerated by the matrix converter625by the vectors of the signals S1to SNto generate signals ΔS1to ΔSNwhose phase is the phase difference between the reception signal and the reference signal. For example, as illustrated inFIG.6, with the phase difference between signal S1and signal SR1denoted by Δθ1and the amplitude of signal S1denoted by r1, the phase of signal ΔS1is Δθ1and the amplitude is r1, as illustrated inFIG.7.FIGS.6and7, as well asFIGS.8through11,FIGS.18and19described later, show the signal S1, etc., on the complex plane. With the real part of the signal S1denoted by I1and the imaginary part denoted by Q1, and the real part of the signal ΔS1denoted by I1′ and the imaginary part denoted by Q1′, I1′ and Q1′ can be obtained according to the following relational expression.

[I1′Q1′]=R1[I1Q1]

Similarly, with phase differences between the signals S2and SR2, between the signals S3and SR3, . . . , and between the signals SNand SRNdenoted by Δθ2to ΔθN, the amplitudes of the signals S2to SNare r2to rN. From real parts I2to IN, imaginary parts Q2to QN, and rotation matrices R2to RNof the signals S2to SN, the real parts I2′ to IN′ and imaginary parts Q2′ to QN′ of the signals ΔS2to ΔSNare calculated. The multiplier627outputs the signals ΔS1to ΔSNindividually to the integrator623.

As illustrated inFIG.5, the integrator623includes a summed signal generator628and an averager629. The signals ΔS1to ΔSNoutput from the multiplier627are input to the summed signal generator628. The summed signal generator628sums the reception signals, thereby performing correlation detection between the reception signal and the reference signal.

When the signals ΔS1to ΔSNare summed, the amplitude increases when the correlation between the reception signal and the reference signal is high, and decreases when the correlation is low. For example, as illustrated inFIG.7andFIG.8, in a case where the phases Δθ1and Δθ2of the signals ΔS1and ΔS2are equal to each other, the amplitude increases when the signal ΔS2is added to the signal ΔS1, as illustrated inFIG.9. On the other hand, as illustrated inFIG.10, in a case where the phase Δθ2of the signal ΔS2is significantly different from the phase Δθ1of the signal ΔS1, the amplitude decreases when the signal ΔS2is added to the signal ΔS1, as illustrated inFIG.11.

Summing the signals ΔS1to ΔSNallows the amplitude of the complex signal generated by the summed signals to represent the level of the correlation between the reception signal and the reference signal. The summed signal generator628outputs the complex signal generated by summation of the signals ΔS1to ΔSNto the averager629.

The averager629generates an averaged complex signal by dividing the amplitude of the output signal from the summed signal generator628by N. The complex signal averaged by the averager629is output to the amplitude converter624.

In an alternative embodiment, the integrator623may be configured to sum only the signals included in a range set by a predefined condition among the signals ΔS1to ΔSN. For example, since the components at both ends of the frequency band in the reception signal have low S/N, setting the summation range such that the corresponding portions are excluded from the reference signal improves the accuracy of code determination.

The amplitude converter624converts the complex signal received from the averager629into an amplitude signal. Specifically, the amplitude converter624calculates the absolute value from the real and imaginary parts of this complex signal and outputs this absolute value as an amplitude. The amplitude signal generated by the amplitude converter624is output to the determiner8as a correlation signal.

As illustrated inFIG.4, the down-chirp filter620B, like the up-chirp filter620A, includes the reference signal holder621, the vector rotator622, the integrator623, and the amplitude converter624. The reference signal holder621through the amplitude converter624of the down-chirp filter620B are configured similarly to the reference signal holder621through the amplitude converter624of the up-chirp filter620A. In the down-chirp filter620B, a complex reference signal generated through quadrature detection of the reference signal corresponding to the down-chirp is input from the reference signal processor7to the reference signal holder621, and correlation detection is performed between the complex reception signal and this complex reference signal. The amplitude signal generated by the amplitude converter624is output to the determiner8as a correlation signal.

The determiner8determines the code included in the reception signal based on the correlation signals output from the matched filter6. The determiner8corresponds to a code determiner. The determiner8makes an object detection determination based on the reception signal and a result of code determination. The determiner8calculates peaks of the up-chirp correlation signal and the down-chirp correlation signal based on the correlation outputs of the up-chirp filter620A and the down-chirp filter620B. The determiner8compares these peaks to determine that the code corresponding to the correlation signal with the larger peak is included in the reception signal, thereby making an object detection determination. The determiner8transmits the result of object detection determination to the controller3.

The controller3is connected to the ultrasonic sensor2via an on-board communication line to enable information communication, and is configured to control the transmit and receive operations of the ultrasonic sensor2. The controller3is provided as a so-called sonar ECU and includes an on-board microcomputer formed of a CPU, a ROM, a RAM, a non-volatile rewritable memory, and other components, which are not shown in the figure. The ECU is an abbreviation for Electronic Control Unit. The non-volatile rewritable memory is, for example, an EEPROM, a Flash ROM or the like. EEPROM is an abbreviation for Electronically Erasable and Programmable Read Only Memory. The ROM, RAM and the like are non-transitory tangible storage media.

The operation of the object detection device1will now be described. The object detection device1repeatedly performs the object detection process, including the process illustrated inFIG.12. In the object detection process, first, a transmission instruction is issued from the controller3to the drive signal generator5, and a probe wave is transmitted from the transducer41based on the drive signal generated by the drive signal generator5. Upon detection of reception of the ultrasonic signal by the transceiver4, the object detection device1performs the process illustrated inFIG.12to detect an object.

First, at step S101, the quadrature detector61quadrature detects the reception signal output from the transceiver4to generate a complex reception signal and outputs it to the correlation filter62. The quadrature detector71quadrature detects the reference signals corresponding to respective ones of the up-chirp and the down-chirp output from the driving signal generator5, generates complex reference signals, and outputs the complex reference signals to the correlation filter62.

Subsequently, at step S102, the correlation filter62performs correlation detection between the complex reception signal output from the quadrature detector61and the complex reference signal corresponding to the up-chirp, and outputs a correlation signal to the determiner8. The correlation filter62performs correlation detection between the complex reception signal and the complex reference signal corresponding to the down-chirp, and outputs a correlation signal to the determiner8.

Subsequently, at step S103, the determiner8determines whether the peak of the correlation signal for the up-chirp output from the correlation filter62is greater than the peak of the correlation signal for the down-chirp. If the peak of the correlation signal for the up-chirp is greater than the peak of the correlation signal for the down-chirp, the determiner8stores the result of code determination that the reception signal includes the up-chirp, and makes an object detection determination based on this result at step S104. If the peak of the correlation signal for the up-chirp is less than or equal to the peak of the correlation signal for the down-chirp, the determiner8stores the result of code determination that the reception signal includes the down-chirp, and makes an object detection determination based on this result.

For example, the determiner8calculates a distance to the object based on a length of time from transmission of the probe wave to when the amplitude of the reception signal whose code matches that of the transmission signal becomes greater than or equal to a predefined value, and transmits a result of calculation to the controller3. Based on the result of calculation of the distance and the speed of the own vehicle and the like, the controller3determines whether a possibility of colliding with the object is high. According to the result of determination, avoidance control or braking control is performed. After completion of step S104or S105, the object detection unit1terminates the process.

Advantages of the present embodiment will be described. As described above, in the present embodiment, the reception signal and the reference signals are converted into complex signals through quadrature detection, and the code included in the reception signal is determined by correlation detection between the complex reception signal and each of the complex reference signals. Converting the reception signal and the reference signals into complex signals in this manner allows for correlation calculation by vector and matrix operations and down-sampling, which can reduce the amount of calculation in the correlation filter62and thus reduce the circuit area.

Second Embodiment

A second embodiment will now be described. This embodiment is different from the first embodiment in that a configuration for normalizing the complex signals is added, and the other parts are similar to those of the first embodiment. Therefore, only the differences from the first embodiment will be described.

As illustrated inFIG.13, the matched filter6of the present embodiment includes a normalizer63and a corrector64, in addition to the quadrature detector61and the correlation filter62. The reference signal processor7includes a normalizer72in addition to the quadrature detector71. The drive signal generator5, the matched filter6, the reference signal processor7, and the determiner8are configured, for example, as a DSP having the functions programmed, such as the above-described drive signal generation, quadrature detection, correlation detection, code determination, object detection determination, and normalization, delay correction, amplitude correction and the like as described later.

The normalizer63normalizes the complex reception signal output from the quadrature detector61such that the amplitude of the complex reception signal is constant. The normalizer63corresponds to a second normalizer. As illustrated inFIG.14, the normalizer63includes an amplitude converter631, a moving average filter632, and a vector normalizer633. The complex reception signal output from quadrature detector61is input to the amplitude converter631and to the vector normalizer633.

The amplitude converter631converts the complex reception signal output from the quadrature detector61into an amplitude. The amplitude converter631calculates the amplitudes r1to rNfrom the real parts I1to INand the imaginary parts Q1to QNfor the signals S1to SN. That is, the amplitude r1is calculated as r1=√(I12+Q12), and the amplitudes r2to rNare calculated in the same manner. The result of amplitude calculation by the amplitude converter631is input to the moving average filter632and the vector normalizer633.

The moving average filter632calculates a moving average of the amplitudes r1to rNand generates an envelope of the amplitude of the complex reception signal. The settings of the moving average filter632are received from the controller3. The envelope of the amplitude of the complex reception signal generated by the moving average filter632is output to the corrector64and to the determiner8. In the present embodiment, the normalizer63includes the moving average filter632. Alternatively, the normalizer63may include an LPF instead of the moving average filter632. In addition, the normalizer63may not include the moving average filter632, and the output of the amplitude converter631may be output directly to the corrector64and to the determiner8.

The vector normalizer633converts, based on the amplitudes r1to rNinput from the amplitude converter unit631, the complex reception signal received from the quadrature detector unit61into a unit vector by normalizing the amplitude while leaving the phase unchanged. Specifically, the vector normalizer633divides the complex reception signal by its original amplitude. That is, the real parts I1to INof the signals S1to SNare converted to I1/r1to IN/rN, and the imaginary parts Q1to QNof the signals S1to SNare converted to Q1/r1to QN/rN. In the present embodiment, the normalized signals S1to SNare input to the correlation filter62. In the multiplier627of the vector rotator622, I1/r1to IN/rNand Q1/r1to QN/rNare used instead of I1to INand Q1to QNto perform the operations according to the relational expression 2.

As described in the present embodiment, the case in which the complex reception signal is normalized such that the amplitude is one has been described. Alternatively, the amplitude of the complex reception signal may be normalized to a value different from one. As described later, in the present embodiment, each complex reference signal is normalized such that the amplitude is one. Alternatively, the amplitude of each complex reference signal may be normalized to a value different from one.

Thus, in the present embodiment, the complex reception signal is normalized by the normalizer63. The normalized complex reception signal is input to the correlation filter62, where correlation detection with each of the complex reference signals is performed. The correlation filter62of the present embodiment outputs the correlation signals to the corrector64.

The normalizer72normalizes each of the complex reference signals output from the quadrature detector71such that the amplitude is constant. The normalizer72corresponds to a first normalizer. The normalizer72has the same configuration as the amplitude converter631and the vector normalizer633of the normalizer63. The complex reference signal normalized by the normalizer72to have an amplitude of one is output to the correlation filter62. Specifically, the up-chirp filter620A of the correlation filter62receives the normalized signals SR1-SRNcorresponding to the up-chirp from the normalizer72. The down-chirp filter620B of the correlation filter62receives the normalized signals SR1-SRNcorresponding to the down-chirp from the normalizer72. In each of the up-chirp filter620A and the down-chirp filter620B, correlation detection is performed between the normalized complex reception signal and the normalized complex reference signal, and the correlation signal is output.

The corrector64corrects the amplitudes and the like of the correlation signals output from the correlation filter62. As illustrated inFIG.15, the corrector64includes delay correctors641and642, and multipliers643and644. The delay correctors641and642delay the phase of the amplitude signal output from the normalizer63according to phase delays of the output signals of the up-chirp filter620A and the down-chirp filter620B. The delay correctors641and642receive the amplitude signal generated by the moving average filter632of the normalizer63, and the amplitude signals delay-corrected by the delay correctors641and642are output to the multipliers643and644, respectively.

Each of the multipliers643and644multiplies the amplitude of the correlation signal output from the corresponding one of the up-chirp filter620A and the down-chirp filter620B of the correlation filter62by the amplitude before normalization and restore it to its original magnitude. This enables the determiner8to make a code determination and an object detection determination by comparing the original amplitude with a predefined threshold value. The amplitude signal whose amplitude has been restored to its original magnitude by each of the multipliers643and644is output to the determiner8.

In the object detection process of the present embodiment, at step S101inFIG.12, after the quadrature detector61converts the reception signal into a complex reception signal, the normalizer63normalizes the complex reception signal output from the quadrature detector61to set the amplitude to one. The complex reference signal output from the quadrature detector71is normalized by the normalizer72to have an amplitude of one. At step S102, the correlation filter62performs correlation detection between the normalized complex reception signal and each of the normalized complex reference signals, and at step S103, the determiner8makes a code determination based on the result of correlation detection.

Advantages of the present embodiment will now be described. The signal width of output of the correlation filter62is inversely proportional to the frequency bandwidth of the reception signal, as illustrated inFIG.16. The broader the frequency bandwidth, the narrower the signal width of the filter output.FIG.16illustrates the signal widths measured by the inventors by varying the frequency bandwidth, and the dashed-dotted line is an approximate curve of the measurements.

Since there are a plurality of reflection points in an obstacle with a complicated shape, it is desirable to narrow the signal width of the filter output in order to achieve a high accuracy of code determination. However, a microphone used as a transducer in an onboard sensor has a narrow band frequency characteristic, as illustrated inFIG.17. That is, when a microphone with such characteristics is used for the transducer41, the transmit and receive sensitivity is high near the resonant frequency f0 that is the resonant frequency of the transducer41, but at frequencies away from the resonant frequency f0, the transmit and receive sensitivity is low.

Thus, for example, when transmitting a chirp signal such that fc=f0, the component at the center frequency fc becomes larger while the components at frequencies away from the center frequency fc become smaller. Only the components at or near the center frequency fc of the entire band can be fully utilized.

Specifically, assuming that, among the signals S1to SNforming the complex reception signal, the signal corresponding to the resonant frequency f0 is SAand the signal corresponding to a frequency away from the resonant frequency f0 is SB, the amplitudes of the signals SAand SBare, for example, as illustrated inFIG.18. That is, the amplitude of the signal SAis greater than one, and the amplitude of the signal SBis less than one.

As illustrated inFIG.16, the hardware limitation of such a microphone makes it difficult to reduce the signal width of the filter output to a desirable value, which may cause an erroneous determination of the code when detecting an obstacle with a complicated shape, such as a vehicle or a fence. As illustrated inFIG.18, large differences in amplitude between signals S1-SNwith frequency may cause the result of correlation detection to be affected by the amplitudes near the resonant frequency f0, leading to an erroneous determination of the code.

In contrast, in the present embodiment, the complex reception signal output from the quadrature detector61is normalized by the normalizer63prior to correlation detection. That is, as illustrated inFIG.19, the amplitudes of the signals S1to SNare made equal to one. InFIG.19, only signals SAand SBare shown among the signals S1to SN. This can reduce the effect on the frequency characteristics of the microphone. As illustrated inFIG.20, the frequency band of the reception signal becomes broader and the signal width after correlation detection becomes narrower, thus enabling suppression of erroneous code determination.

FIG.20illustrates changes in the frequency band arising from normalization. InFIG.20, the dashed-dotted line indicates the amplitude of the complex reception signal generated by the quadrature detector61, and the solid line indicates the amplitude of the complex reception signal normalized by the normalizer63. InFIG.20, fLPF is the cut-off frequency of the LPF612.

Thus, the effect on the frequency characteristics of the microphone is decreased by the normalizer63normalizing the reception signal that varies in amplitude due to the frequency characteristics of the transducer41.

For example, when a probe wave is transmitted toward an object with a complicated shape, such as a fence, a plurality of reflected waves may be returned. If correlation detection is performed without normalization of the complex reception signal, the peak of the correlation output will be lower than the amplitude signal of the original probe wave indicated by the dashed-dotted line, as illustrated inFIG.21. For example, setting the threshold for reflected wave detection as indicated by the dashed line results in a larger signal width and reduced resolution due to interference from reflected waves.

In contrast, normalization of the complex reception signal suppresses reduction of the peak of the correlation output, reduces the signal width, and suppresses the reduction of resolution due to interference from reflected waves, as illustrated inFIG.22.

As described above, in the present embodiment, the complex reception signal is normalized prior to correlation detection, which reduces the effect on the frequency characteristics of the transducer41. This can suppress reduction of the peak of the correlation output and can improve the resolution by decreasing the signal width, which improves the accuracy of code determination. In addition, normalizing the amplitude of the complex signal to one allows the operations to be simplified by a known formula. Normalization of the complex reference signal prior to correlation detection can further reduce the effect on the frequency characteristics of the transducer41. Normalization of the complex reference signals allows them to be treated as trigonometric functions, which can facilitate conversion to rotation matrices, thereby further reducing the amount of calculation.

Third Embodiment

A third embodiment will now be described. This embodiment is different from the second embodiment in that a configuration for rotating the phases of the complex signals is added, and the other parts are similar to those of the second embodiment. Therefore, only the differences from the second embodiment will be described.

The object detection device1of the present embodiment has a configuration to further broadens the frequency band of the reception signal by phase rotation. As illustrated inFIG.23, the matched filter6of the present embodiment includes a phase rotator65, in addition to the quadrature detector61, the correlation filter62, the normalizer63, and the corrector64. The reference signal processor7includes a phase rotator73, in addition to the quadrature detector71and the normalizer72. The drive signal generator5, the matched filter6, the reference signal processor7, and the determiner8are configured, for example, as a DSP having the functions programmed, such as the above-described drive signal generation, quadrature detection, normalization, correlation detection, delay correction, amplitude correction, code determination, object detection determination, and phase rotation and the like as described later.

The phase rotator65rotates the phase of the complex reception signal. The phase rotator65corresponds to a first phase rotator. The phase rotator65receives the complex reception signal normalized by the normalizer63. The complex reception signal whose phase is rotated by the phase rotator65is output to the correlation filter62.

The phase rotator65specifically processes the received signal as follows. That is, using I′=cos θ, Q′=sin θ, cos 2θ=1−2 sin2θ, and sin 2θ=2 sin θ cos θ, cos 2θ and sin 2θ are calculated from I′ and Q′, where I′ is the real part of the normalized complex reception signal, Q′ is the imaginary part of the normalized complex reception signal, and θ is the phase. The real and imaginary parts of the new complex reception signal are output as cos 2θ and sin 2θ, respectively.

The phase rotator73rotates the phase of each of the complex reference signals. The phase rotator73corresponds to a second phase rotator. The phase rotator73receives the complex reference signal normalized by the normalizer72, and the complex reference signal whose phase is rotated by the phase rotator73is output to the correlation filter62. In the phase rotator73, phase rotation is performed in the same manner as in the phase rotator65. The correlation filter62performs correlation detection between the phase-rotated complex reception signal and each of the phase-rotated complex reference signals, and outputs correlation signals.

FIG.24illustrates changes in the frequency band arising from phase rotation. InFIG.24, the dashed-dotted line indicates the amplitude of the complex reception signal normalized by the normalizer63, and the solid line indicates the amplitude of the complex reception signal phase rotated by the phase rotator65. As described in the second embodiment, normalization broadens the frequency band while the phase rotation further broadens the apparent frequency band, as illustrated inFIG.24.

The amount of phase rotation is equal to an integral multiple, e.g., twice as above. Alternatively, the phase may be rotated by other multiples. For example, in the phase rotators65and73, a twice the phase rotation may be performed twice and a signal with four times the phase rotated may be output, such as cos 4θ=1−2 sin22θ and sin 4θ=2 sin 2θ cos 2θ.

The amount of phase rotation may be changed according to a predefined condition. For example, the higher the magnification ratio, the broader the frequency band after phase rotation. However, since the effect of Doppler shift increases, the magnification ratio may be lower than the predefined value when traveling straight ahead at a high vehicle speed and higher than the predefined value when reversing at a low vehicle speed.

In the object detection process of the present embodiment, at step S101inFIG.12, the quadrature detector61converts the reception signal into a complex signal, the normalizer63normalizes the complex reception signal, and then the phase rotator65performs phase rotation by rotating the phase of the normalized complex reception signal. After the normalizer72normalizes each complex reference signal output from quadrature detector71, the phase rotator73performs phase rotation by rotating the phase of each normalized complex reference signal. At step S102, the correlation filter62performs correlation detection between the phase-rotated complex reception signal and each of the phase-rotated complex reference signals, and at step S103, the determiner8makes a code determination based on this correlation detection result.

Advantages of the present embodiment will now be described.FIGS.25and26illustrate results of experiments conducted by the inventors, showing outputs of the correlation filter62when a probe wave was transmitted toward a pole with a diameter of 60 mm. In these experiments, chirp signals with a narrower frequency modulation range than the chirp signals in the probe waves were used as the reference signals. Specifically, the lower and upper limits of the frequency modulation range of the probe wave were set to f1 and f2, respectively, and the lower and upper limits of the frequency modulation range of the reference signals were set to f3 and f4, respectively, such that f1<f3<f4<f2.

FIG.25illustrates the output of the up-chirp filter620A when a probe wave that includes an up-chirp signal is transmitted, andFIG.26illustrates the output of the down-chirp filter620B when a probe wave that includes a down-chirp signal is transmitted. In each ofFIGS.25and26, the solid line indicates the correlation output when phase rotation is performed, the dashed-dotted line indicates the correlation output when phase rotation is not performed, and the dashed line indicates a threshold for reflected wave detection. As illustrated inFIGS.25and26, the phase rotation reduces the signal widths of the output signals of the correlation filter62.

Each ofFIG.27andFIG.28illustrates the output of the correlation filter62when a probe wave including a down-chirp signal was transmitted wave toward a mesh fence.FIG.27is an example of output signals of the correlation filter62when no phase rotation was performed, andFIG.28is an example of output signals of the correlation filter62when phase rotation was performed. In each ofFIG.27andFIG.28, the solid line indicates the output signal of the down chirp filter620B, and the dashed-dotted line indicates the output signal of the up-chirp filter620A.

InFIG.27, the output peak of the up-chirp filter620A is greater than the output peak of the down-chirp filter620B. Thus, it is erroneously determined that the reception signal includes an up-chirp signal. On the other hand, inFIG.28, the output peak of the down-chirp filter620B is greater than the output peak of the up-chirp filter620A. Therefore, it is correctly determined that the reception signal includes a down-chirp signal.

Without phase rotation, there is a risk of occurrence of erroneous determinations as described above, while with phase rotation, the output signals of the up-chirp filter620A and down-chirp filter620B are less dull, as illustrated inFIG.28, resulting in fewer erroneous determinations. In experiments conducted by the inventors, addition of the phase rotation process improved the code recognition rate from 88% to 95% when probe waves were transmitted toward the mesh fence.

As described above, in the present embodiment, phase rotation of the complex signals leads to more reduced signal widths and improves the accuracy of code determination. Since the complex reception signal and the complex reference signals are normalized prior to phase rotation, the phase rotation process may be performed using the double-angle formulae as described above, thereby reducing the amount of calculation for phase rotation.

OTHER EMBODIMENTS

The present disclosure should not be limited to the embodiments described above and can be modified as deemed appropriate. Needless to say, in the above-described embodiments, the components of the embodiments should not be necessarily deemed to be essential unless explicitly described or they are fundamentally and obviously essential, for example.

For example, as illustrated inFIG.29, each of the probe wave and reference signals may be formed of a component of one frequency lower than the resonant frequency f0 of the transducer41and a component of one frequency higher than the resonant frequency. Using a probe wave of a frequency different from the resonant frequency f0 of the transducer41in this manner can reduce the effects on the frequency characteristics of the transducer41and improve robustness of the object detection device1. InFIG.29, the frequency of the probe wave is modulated from a frequency lower than the resonant frequency f0 to a frequency higher than the resonant frequency f0. Alternatively, the frequency of the probe wave may be modulated from a frequency higher than the resonant frequency f0 to a frequency lower than the resonant frequency f0.

In the second embodiment, the complex reception signal and the complex reference signals are all normalized. In an alternative embodiment, only the complex reference signals may be normalized, without normalizing the complex reception signal. In an alternative embodiment, only the complex reception signal may be normalized without normalizing the complex reference signals.

In an alternative embodiment to the above third embodiment, the normalized and phase-rotated complex reception signal may be restored to its original amplitude before performing correlation detection. For example, as illustrated inFIG.30, the matched filter6includes an amplitude multiplier66. The complex reception signal that has been phase-rotated by the phase rotator65and the amplitude calculated by the normalizer63are input to the amplitude multiplier66. The amplitude multiplier66multiplies the complex reception signal by the amplitude before normalization to restore the amplitude of the complex reception signal, and outputs the amplitude restored complex reception signal to the correlation filter62. In such an embodiment, correction of the amplitude by the corrector64is not needed. Similarly, the normalized complex reference signals may be restored to their original amplitudes before performing correlation detection.

In an alternative embodiment, the phase rotators65and73may be added to the first embodiment to perform phase rotation of the complex reception signal and the complex reference signals output from the quadrature detectors61and71.

In the above-described embodiments and modifications, the drive signal generator, the matched filter, the reference signal processor, the determiner, the controller and the like and the method of them described in the present disclosure may be implemented by a dedicated computer including a processor and a memory programmed to execute one or more functions embodied by computer programs. Alternatively, the drive signal generator, the matched filter, the reference signal processor, the determiner, the controller and the like and the method of them described in the present disclosure may be implemented by a dedicated computer including a processor formed of one or more dedicated hardware logic circuits, or may be implemented by one or more dedicated computers including a combination of a processor and a memory programmed to execute one or more functions and a processor formed of one or more dedicated hardware logic circuits. The computer programs may be stored, as instructions to be executed by a computer, in a non-transitory, tangible computer-readable storage medium.