Patent ID: 12235365

DETAILED DESCRIPTION OF THE INVENTION

FIG.1is an overview of one embodiment of the system depicting an eight element sCRPA11broadband receive antenna array integrated with an Iridium TX transceive patch antenna element12. (The transceive patch element12supports Iridium user uplink data transmissions to the Iridium SV.) The specific types of patch element and array, and number of antenna elements are illustrative and not limiting. The antenna may be composed of multi-element arrays of patch-elements, and/or polarimetric elements, such as bent monopole elements and/or loop elements. Some, not all, classes of polarimetric-element arrays require mode-forming. For example, mode-forming is required with antenna systems based on small CRPA (Controlled Reception Pattern Antenna) arrays, whereas the larger ADAP (Advanced Digital Antenna Production) antenna array does not require mode-forming for effective antijam processing of the received signals. All such antenna array based implementations requiring antijam processing with and without mode-forming are within the scope of this disclosure and the claims herein.

Signal outputs from the antenna array are downconverted and digitized into complex baseband samples, which are optionally available for digital modeforming (e.g., Butler array), if desired, prior to the signal processing for suppressing the GPS-bands jamming and Iridium-band jamming13. The depicted embodiment shows an Iridium unprotected bypass mode of operation14, whereby, if desired by the user, the Iridium radio bypasses its AJ protection. A circulator15enables passage of both Iridium uplink and downlink signals.

The GPS/Iridium AJ system13implements Nulling for the protection of the GPS L1 and/or L2 band transmissions and Beamforming for the protection of the Iridium user downlink transmission. Both Nulling and Beamforming algorithms run concurrently. Algorithms may be implemented in software (SW) or in a combination of hardware (HW), such as Field Programmable Gate Array (FPGA) or Application Specific Integrated Circuit (ASIC) technology, and software running on a general purpose processor (GPP). An illustrative split between HW and SW functionality appears inFIG.2. The most compute-intensive aspects of the algorithms (e.g., fast-Fourier transform (FFT), matrix inversion/factorization) are preferably implemented in HW (but may also be implemented in SW), while less demanding aspects are implemented in SW (e.g., determination of beam direction, HW control). The particular division of work between HW and SW described herein is understood to be illustrative and not limiting.

The Nulling solution for the protection of GPS involves computing sample covariance matrices from the complex-valued, baseband (and, optionally, mode-formed) data output from the antenna array. The inverse of the sample covariance matrix is multiplied by a pseudo-steering vector to obtain the STAP-filter's complex-valued weight vector. The pseudo-steering vector is a vector with a unity-valued element that normally corresponds to the branch center-tap of the antenna array reference element (or the reference omni-mode for mode-formed arrays). The Nulling solution is well-known art with already known performance characteristics, so further details are not provided. See, e.g.,Understanding GPS: Principles and Applications, Editors: E. D. Kaplan and C. J. Hegarty, Artech House, 2006.

The Beamforming solution is closely examined inFIG.2with an illustrative but not limiting embodiment. Digital down converters (DDCs)21convert the real-valued outputs of analog-to-digital (A/D) converters (not shown) to complex baseband form. The example here shows without limitation that the eight DDC converter-bank outputs (which correspond, respectively, to the eight signals received from the eight antenna elements) are, optionally, input to a digital mode-forming block22, if mode forming is implemented, to select the five most efficient modes for the example sCRPA array depicted inFIG.1. (The choice of five modes is illustrative, and not limiting.) Alternatively, the mode-forming step may be avoided and further STAP filtering performed directly using the complex baseband signals. The complex-valued time-domain samples (from mode forming, if implemented, or otherwise) are also input to the aforementioned Nulling (not shown). However, for use in Beamforming the samples are preferably buffered (FIFO23) before being STAP-filtered. The FIFOs permit matching the complex baseband samples that are to be filtered to their appropriate filter weights. The weights themselves are obtained by processing K-sample buffered samples24(e.g., K=8192 complex samples/mode).

In this latter processing, described in further detail below, the time-domain K-samples are Fourier transformed (e.g., via NF-bin fast Fourier transform (FFT)), and the transformed data is formed into cross-spectral density matrices (CSDMs)24, which may also be referred to as Fourier domain sample covariance matrices (per Owsley). The CSDMs are employed in the computation of Fourier domain STAP filter weights25as discussed below. Inverse FFTs are employed to convert the Fourier domain weights into the time-domain25for use in the STAP filter26. The STAP filter employs NFtaps per finite impulse response (FIR) filter branch (FIG.3) where one filter branch is allocated to each antenna array element or mode from the antenna array output. (For illustration only. inFIG.3, L=5 for a 5-mode system.) The complex-valued samples are digitally up-converted (See e.g., DUC27), which is the inverse processing to that of the DDC, followed, if and as required, by digital-to-analog (D/A) conversion28and RF up-conversion (not shown), and onward transmission to the Iridium radio.

FIG.3illustrates the FIR filter signal processing31within the Beamformer26, where NF-tap filtering is applied to each of the L modes or antenna element signals (inFIG.2we illustrate for L=5 modes as an example). The term wk,ninFIG.3denotes the tap n complex-valued weight applied to the antenna element k or mode-k branch. The antenna element k or mode-k antenna array response at the frequency ω in the Iridium space vehicle (SV) signal direction (θ, φ) (where θ=azimuth, φ=elevation in the platform's local Cartesian frame), may be represented as Ak(θ, φ, ω). We call (θ, φ) the Iridium signal direction-of-arrival (DOA).

Thus,FIG.3also indicates that beamforming requires antenna pattern data (inclusive of RF electronics effects) for obtaining the correct steering vector to the Iridium SV. The direction-of-arrival (DOA) depends on the line-of-sight (LOS) to the Iridium SV from the platform hosting the antijam system (AJ system), inertial navigation system (INS), and GPS and Iridium receivers. The LOS is determined from both platform orientation ((INS)-derived information) and platform position (GPS-derived information) in space. For satellite communication, Iridium SV almanac information is also needed to determine which Iridium SV is in view, as LOS also requires position information about the Iridium SV.

The Beamforming algorithm employed herein is Capon's algorithm (See, e.g., Capon, and Owsley). This algorithm minimizes the STAP filter output power to achieve nulling of the interference, while at the same time the gain response of the STAP filter is constrained to unity to prevent nulling of the Iridium user's downlink signal.

FIG.4depicts the key signal processing steps involved in obtaining the Beamformer weights. The DDC21sample sequences are denoted xk(nT) where T is the DDC sampling interval and k=1,2, . . . , L−1, L for an antenna array of L antenna elements or modes, (e.g., L=5). InFIG.4, sample blocks of length K complex samples per mode or antenna element are sub-divided into non-overlapping segments each of length NFsamples, and thus K=MNF, for which M is the number of segments. Each NFlength segment is transformed using NF-bin DFT/FFT operations to obtain a K-sample transformed data (Fourier coefficients) set per mode. Relatively long sample blocks (large K) are needed to obtain good covariance estimates. It has been found in experimentation with a prototype system that K may range from 2048 to 8192 samples. Prototyping also shows that NF=16 yields acceptable performance.

The Fourier bin m segment (or block) v FFT coefficients for each antenna element or mode are accumulated into a length L complex-valued column vector denotedXm,v=[X1,v(ωm) X2,v(ωm) . . . XL,v(ω)]TinFIG.4. The Fourier bin m cross-spectral density matrix (CSDM) (also called Fourier domain sample covariance matrix) is obtained via the outer products summations:

Rm=1M⁢∑v-0M⁢1⁢X¯m,v⁢X¯m,vH∈CL×L(1)
These CSDMs are applicable for the observation time k, and the computed beamformer STAP weights (see below) apply to these same samples as well.

Following Owsley, the following quadratic programming problem (QPP), when solved using the steering vectoram,p(defined below), yields the Fourier domain STAP weights,Ŵm:

W^¯m=argminW¯m⁢W_mH⁢Rm⁢W¯m⁢⁢subject⁢⁢to⁢⁢W_mH⁢a_m,p=1(2⁢a)
for m=0,1, . . . , NF−1 in general, or only some relevant subset of these indices such as those covering the Iridium user downlink band.

The steering vector is
am,p=[A1(θp,φp,ωm)A2(θp,φp,ωm) . . .AL(θp,φp,ωm)]T∈CL(2b)
This is from pre-stored antenna pattern and includes all RF electronics effects at frequency ωm(baseband equivalent, normalized frequency scale) in the pth signal direction (θp,φp) (i.e., azimuth, elevation) over all of the available antenna array modes k=1,2, . . . L. Thus, Ak(θ, φ, ω) represents the mode (or antenna element, if there is no mode forming) k antenna pattern.

The linear equality constraint in (2a) constrains the STAP filter gain to be unity in the desired DOA, and across the band for which the constraints are applied. Implicitly, a linear phase response is also imposed over that band. However, the Iridium transmissions are very narrowband and so the linear phase response characteristic of the solution from solving (2a) is not critical.

From the method of Lagrange multipliers, the solution to the constrained QPP in (2a) is as follows, which is the well-known solution to the Capon beamforming weights design problem:

W^¯m=1a¯m,pH⁢Rm1⁢a¯m,p⁢Rm1⁢a¯m,p=[W^1⁡(ωm)W^2⁡(ωm)⋮W^L⁡(ωm)]∈CL(3)

The computation (3) is done for bin m covering the 10.5 MHz Iridium user band at complex baseband. The Fourier domain weights for a bin that is not in this band are simply set to zero. From Equation (3) we see that a given L-element solution vector contains the Fourier domain weights for all the modes or antenna elements at that bin index. Thus, an implementation must search through all solution vectors to extract all of the Fourier domain weights for any given mode or antenna element of interest.

The Fourier domain weights from (3) are transformed into the time-domain using the inverse FFT so that filtering may be realized using STAP.

The Beamformer design described herein has been implemented as a prototype and tested a number of times against live in-sky Iridium satellite signals with broadband (BB) injected jamming signals. The Beamforming anti-jam (AJ) test procedure used was as follows:a. Align 0-deg azimuth (AZ) angle of the antenna array to the true North direction.b. Use software to predict a tracked Iridium satellite and ensure it is a rising satellite.c. Without jamming, let the Iridium phone track the satellite by going to “registered” state.d. Turn on the broadband (BB) jammer to jam out the Iridium phone. The Iridium phone should go to “searching for network” state after the jammer is turned on. The jamming power at this time is the OMNI break point. Because we don't have enough time to dwell in the OMNI mode to search for the OMNI break point during AJ tests, this OMNI break point needs to be determined before the AJ tests.e. Switch from OMNI mode to Beamforming ADAPT mode and wait for the Iridium phone to recover to the “registered” state. (The ADAPT mode pertains to Iridium handset operation in the presence of jammer signals.f. Check if the Iridium phone remains in “registered” state for 15 seconds. If so, increase the jamming power by 5 dB. If the Iridium phone remains in “registered” state for another 15 seconds, then increase the jamming power by another 5 dB. Repeat the process until the Iridium phone goes to “searching for network” state; the jamming power at this time is at the ADAPT break point.g. Record the jamming power at OMNI break point and ADAPT break point, and the difference is the AJ protection of Beamforming.

FIG.5,FIG.6, andFIG.7provide tables of protection levels achieved based on this test procedure in the case of one BB jammer where each test was performed on a different day. Protection levels ranging from 25 to 40 dB were observed over a range of conditions. The OMNI and ADAPT break points are given as attenuation levels in decibels for the respective jammer power.

The scope of the invention is to be limited only by the claims, and not by the drawings or description herein. The words “including”, “comprising”, “having,” “with” and other like words used herein are to be interpreted broadly and comprehensively. Moreover, any embodiments disclosed in the subject application must not be taken as the only possible embodiments. Other embodiments will occur to those skilled in the art and are within the scope of the claims herein.