Abstract:
A frequency synthesizer architecture naturally combines transmitter modulation capability with a wideband all-digital PLL modulation scheme to maximize a digitally-intensive implementation by operating in a synchronous phase-domain. Synchronous logic is provided across a digitally controlled VCO and is synchronous to the VCO output clock by implementing a timing adjustment in association with a reference calculation to allow a frequency control word to contain both channel information and transmit modulation information.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     This invention relates generally to frequency synthesizers, and more particularly to an all-digital phase-domain phase-lock loop (PLL) frequency synthesizer that operates in a synchronous phase-domain to maximize a digitally-intensive architecture. 
     2. Description of the Prior Art 
     Frequency synthesizers using analog circuit techniques are well known in the art. Conventional RF frequency synthesizer architectures are analog-intensive and generally require a low loop bandwidth to reduce the familiar and well-known reference or compare frequency spurs. Low loop bandwidths are acceptable for RF-BiCMOS and RF-SiGe processes with weak digital capabilities. 
     Modern deep sub-micron CMOS processes and their RF-CMOS derivatives, however, are not very compatible with frequency synthesizer designs using analog circuit techniques. The conventional PLL-based frequency synthesizers generally comprise analog-intensive circuitry that does not work very well in a voltage-headroom-constrained aggressive CMOS environment. Such frequency synthesizers do not take advantage of recently developed high density digital gate technology. 
     Newer frequency synthesizer architectures have used sigma-delta modulated frequency divider techniques to randomize the above discussed frequency spurs by randomizing the spurious content at the cost of increased noise floor. These techniques have not significantly reduced the undesirable analog content. Other frequency synthesizer architectures have used direct digital synthesis (DDS) techniques that do not work at RF frequencies without a frequency conversion mechanism requiring an analog solution. Further, previous all-digital PLL architectures rely on an over-sampling clock. Such architectures cannot be used at RF frequencies. 
     In view of the foregoing, it is highly desirable to have a digitally-intensive frequency synthesizer architecture that is compatible with modern CMOS technology. 
     SUMMARY OF THE INVENTION 
     The present invention is directed to an all-digital phase-domain PLL frequency synthesizer that is compatible with deep sub-micron CMOS processes. The all-digital phase-domain PLL frequency synthesizer accommodates direct frequency/phase modulation transmission to remove the requirement for an additional transmitting modulator normally associated with wireless digital transmitters. This is accomplished by operating the PLL entirely in the phase-domain with maximum digital processing content such that the loop can be of high-bandwidth of “type 1” without the need for a loop filter. A “type 1” filter, as used herein, means a loop filter having only one integrating pole in the feedback loop. Only one integrating pole exists due to the VCO frequency-to-phase conversion. It is possible therefore, to eliminate a low-pass filter between the phase detector and the oscillator tuning input, resulting in a high bandwidth and fast response of the PLL loop. 
     According to one embodiment, the all-digital phase-domain PLL frequency synthesizer contains only one major analog component, a digitally-controlled 2.4 GHz voltage controlled oscillator (VCO or dVCO). The PLL loop is an all-digital phase-domain architecture whose purpose is to generate the 2.4 GHz high frequency f osc  for the “BLUETOOTH” standard. The underlying frequency stability of the system is derived from a reference crystal oscillator, such as a 13 MHz TCXO for the global system for mobile communications (GSM) system. The phase of the VCO output is obtained by accumulating the number of significant (rising or falling) edge clock transitions. The phase of the reference oscillator is obtained by accumulating a frequency control word on every significant (rising or falling) edge of the reference oscillator output that is re-clocked via the VCO output. As used herein, “significant edge” means either a “rising” or a “falling” edge. A ceiling element continuously adjusts a reference phase value associated with the accumulated frequency control word by rounding off to the next integer (alternatively, truncating fractional bits necessary) to compensate for fractional-period delays caused by re-clocking of the reference oscillator by the VCO output. The phase error signal is then easily obtained by using a simple arithmetic subtraction of the VCO phase from the adjusted reference phase on every significant edge of the re-clocked reference oscillator output. The phase error signal can then be used as the tuning input to the digitally-controlled VCO directly via a gain element associated with the PLL loop operation. 
     In one aspect of the invention, an all-digital phase-domain PLL frequency synthesizer is provided that allows fast design turn-around using automated CAD tools. 
     In still another aspect of the invention, an all-digital phase-domain PLL frequency synthesizer is provided that achieves much less undesirable parameter variability than normally associated with analog circuits. 
     In yet another aspect of the invention, an all-digital phase-domain PLL frequency synthesizer is provided that allows ease of testability. 
     In yet another aspect of the invention, an all-digital phase-domain PLL frequency synthesizer is provided that requires desirably low silicon area to physically implement. 
     In yet another aspect of the invention, an all-digital phase-domain PLL frequency synthesizer is provided that requires lower power than conventional frequency synthesizers. 
     In still another aspect of the invention, an all-digital phase-domain PLL frequency synthesizer is provided that has direct frequency/phase modulation transmission capability to minimize system transmitter requirements. 
     In still another aspect of the invention, an all-digital phase-domain PLL frequency synthesizer is provided that accommodates the “BLUETOOTH” communication protocol. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     Other aspects and features of the present invention and many of the attendant advantages of the present invention will be readily appreciated as the same become better understood by reference to the following detailed description when considered in connection with the accompanying drawings in which like reference numerals designate like parts throughout the figures thereof and wherein: 
     FIG. 1 illustrates an all-digital PLL synthesizer architecture according to one embodiment of the present invention; 
     FIG. 2 is a simple block diagram illustrating a quantization scheme for fractional-phase detection associated with the synthesizer depicted in FIG. 1; 
     FIG. 3 is a timing diagram illustrating a frequency reference clock signal and a VCO signal for a positive fractional-phase; and 
     FIG. 4 is a timing diagram illustrating a frequency reference clock signal and a VCO signal for a negative fractional-phase. 
     While the above-identified drawing figures set forth alternative embodiments, other embodiments of the present invention are also contemplated, as noted in the discussion. In all cases, this disclosure presents illustrated embodiments of the present invention by way of representation and not limitation. Numerous other modifications and embodiments can be devised by those skilled in the art which fall within the scope and spirit of the principles of this invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     FIG. 1 illustrates an all-digital PLL synthesizer  100  architecture according to one embodiment of the present invention. The synthesizer  100  naturally combines transmitter frequency modulation capability with a wideband, all-digital PLL modulation technique to maximize digitally-intensive implementation by operating in a synchronous phase-domain. The PLL loop is an all-digital phase-domain architecture capable of generating the 2.4 GHz high frequency f osc  for the “BLUETOOTH” standard band. Accordingly, the all-digital phase-domain PLL frequency synthesizer  100  depicted in FIG. 1 contains only one major analog/RF component, a digitally-controlled 2.4 GHz voltage controlled oscillator (dVCO)  104 , being a portion of a numerically-controlled oscillator (NCO)  103 , that also comprises a gain element  105 . The underlying frequency stability of the synthesizer  100  is derived from a frequency reference crystal oscillator  110 , such as a 13 MHz TCXO for the GSM system. 
     The phase θ v (iT v ) of the dVCO  104  clock signal, CKV  114 , with period T v , at time instances iT v , where i is an integer, is obtained by accumulating the number of rising- or falling-edge clock transitions generated via a sinusoidal-to-digital converter  106 .                  θ   v          (     iT   v     )       =       ∑     t   =   0       i   ·     T   v                  f   v          (   t   )                       (     ×   2        π   ·   rad       )                 (   1   )                                
     Without use of frequency reference retiming (described herein below), the phase θ r (kT r ) of a frequency reference clock, FREF, provided by the reference crystal oscillator (FREF)  110 , with period T r , at time instances kT r  where k is another integer, is obtained by accumulating  102  the frequency control word (FCW  116 ) on every rising (or falling) edge of the frequency reference clock FREF. 
     
       
         θ r ( kT   r )= FCW·k·T   r (×2π·rad)  (2) 
       
     
     The PLL operation achieves, in a steady-state condition, a zero averaged phase difference between the dVCO  104  θ v (iT v ) and the reference crystal oscillator  110  θ r (kT r ) phases. Equation (3) below shows the clock period relationship in the mean sense. 
     
       
           FCW=N   i   =N   f   =T   r   /{overscore (T)}   v .   (3) 
       
     
     The present invention is not so limited however, and it shall be readily understood that FCW  116  can be comprised of only an integer or an integer (N i ) and fractional (N f ) parts. 
     As stated herein before, there is no need for a frequency detection function within the phase detector when operating the PLL loop in the phase-domain. This feature importantly allows “type 1” operation of the PLL, where it is possible to eliminate a low-pass filter between the phase detector and the oscillator (dVCO  104 ), resulting in a high-bandwidth and fast response of the PLL loop. 
     The dVCO  104  and the reference crystal oscillator  110  clock domains are entirely asynchronous, making it difficult to physically compare the two digital phase values θ v (iT v ) and θ r (kT r ) at different time instances iT v  and kT r . Mathematically, θ v (iT v ) and θ r (kT r ) are discrete-time signals with incompatible sampling times and cannot be directly compared without some sort of interpolation. The present inventors recognized therefore, it is imperative that any digital-word comparison be performed in the same clock domain. This function is achieved by over-sampling the FREF reference oscillator  110  by the high-rate dVCO  104  output CKV  114 , and using the resulting frequency reference clock CKR  112  to accumulate via accumulator  102  the reference phase θ r (kT r ) as well as to synchronously sample, via latch/register  120 , the high-rate dVCO  104  phase θ v (iT v ). Since the foregoing phase comparison is performed synchronously at the rising edge of CKR  112 , equations (1) and (2) can now be rewritten as follows:                  θ   v          (   k   )       =       ∑     t   =   0       k   ·     T   r                  f   v          (   t   )                       (     ×   2        π   ·   rad       )                 (   4   )                               θ r ( k )= FCW·k·T   r +ε( k )(×2π·rad)  (5) 
     where the index k is the kth transition of the re-timed reference clock CKR  112  and contains an integer number of CKV  114  clock transitions; and ε(k) is the integer-loop quantization error, in the range of ε∈(0,1), that could be further corrected by other means, such as a fractional phase detector  200  discussed in more detail herein below with reference to FIGS. 2-4. 
     In view of the above, the integer phase detector in the synchronous digital phase environment can now be realized as a simple arithmetic subtraction via combinatorial element  122  of the dVCO  104  phase from the reference phase performed every rising edge of the CKR clock  112 . 
     
       
         θ d ( k )=θ r ( k )−θ v ( k )  (6) 
       
     
     The reference re-timing operation can be recognized as a quantization in the dVCO  104  CKV  114  clock transitions integer domain, where each CKV  114  clock transition rising edge is the next integer. Since the synthesizer  100  must be time-causal, quantization to the next CKV  114  clock transition rising edge (next integer), rather than the closest transition (rounding-off to the closest integer), can only be realistically performed. This limitation is then compensated for in the phase-domain by the ceiling element  108  associated with the reference phase since the reference phase θ r (k) is generally a fixed-point arithmetic signal having a sufficiently large fractional part to achieve the required frequency resolution as set forth in Equation 3 above. As stated herein before, a ceiling element  108  continuously adjusts a reference phase value associated with the accumulated frequency control word by rounding to the next integer (alternatively, truncating the fractional bits), thereby compensating for delays caused by re-clocking of the reference oscillator  110  by the VCO output CKV  114 . The ceiling operation (demonstrated via Equation 7) could be easily implemented by discarding the fractional bits and incrementing the integer bits. This technique, however, improperly handles the case when the fractional part is zero, but has no practical consequences. Those skilled in the art will appreciate that this truncation process achieves a timing correction since phase is a characteristic that can be used to describe a time progression. The phase resolution, however, cannot be better than +/−π radians of the dVCO  104  clock, even though the foregoing integer-loop quantization error ε due to reference phase retiming illustrated by Equation 5 is compensated by next-integer rounding operation (ceiling) of the reference phase. 
     
       
         {tilde over (θ)} r ( k )=┌θ r ( k )┐  (7) 
       
     
     FIG. 2 is a simple block diagram illustrating a digital fractional phase detector system  200  capable of accommodating a quantization scheme to measure fractional (sub-Tv) delay differences between the significant edge of the dVCO  104  clock CKV  114  and the FREF oscillator  110  reference clock  112  using a time-to-digital converter (TDC)  201  with a resolution of Δt ref  and express the time difference as a digital word for the synthesizer  100  shown in FIG. 1 according to one embodiment of the present invention. Due to the dVCO  104  edge counting nature of the PLL, it can be appreciated that the phase quantization resolution cannot be better than +/−π radians as stated above. A much finer phase resolution however, is required for wireless applications. Such finer resolution must be achieved without forsaking the requisite digital signal processing capabilities. The solution illustrated in FIG. 2 measures the one-sided fractional (sub-T v ) delay difference between the dVCO  104  clock CKV  114  and the FREF oscillator  110  clock  112  to express the time difference as a digital word ε  202 . According to one embodiment, the maximum achievable timing resolution of the digital fractional phase detector  200  is determined by an inverter delay associated with a given CMOS process, and is about 40 psec for the C035.1 CMOS process developed by Texas Instruments Incorporated of Dallas, Tex. The digital fractional phase is determined by passing the dVCO  104  clock CKV  114  through a chain of inverters (not shown), such that each inverter output would produce a clock pulse slightly delayed from that of the immediately previous inverter. The resultant staggered clock phases would then be sampled by the same reference clock. 
     As seen in FIGS. 3 and 4, position of the detected transition from 0 to 1 would indicate a quantized time delay ΔT r  between the FREF  110  sampling edge and the rising edge  302  of the dVCO clock, CKV  114  in Δt res  multiples; and position of the detected transition from 1 to 0 would indicate a quantized time delay ΔT f  between the FREF  110  sampling edge and the falling edge  400  of the dVCO clock, CKV  114 . Because of the time-causal nature of the foregoing digital fractional phase detection process, both time delay values ΔT r  and ΔT f  must be interpreted as positive. This is fine if ΔT r  is smaller than ΔT f  since this situation corresponds to the negative phase error of the classical PLL loop in which the VCO edge is ahead of the reference edge and, therefore, the phase sign has to be negated. If ΔT r  is greater than ΔT f  however, the situation becomes problematic since the situation now corresponds to the positive phase error of the classical PLL loop. The time lag between the reference edge FREF  110  and the following rising edge of CKV  114  must be based on the available information regarding the delay between the preceding rising edge of CKV  114  and the reference edge FREF  110  as well as the clock half-period which can be expressed as a difference as shown by Equation 8 below.                  T   v     /   2     =     {             Δ                   t   r       -     Δ                   t   r                 Δ                   t   r       ≥     Δ                   t   f                     Δ                   t   f       -     Δ                   t   r             otherwise                   (   8   )                                
     The foregoing analysis is summarized in Equation 9 below, where Δt frac  is the digital fractional phase detector error.                Δ                   t   frac       =     {             -   Δ                     t   r               Δ                 t     ≤     Δ                   t   f                     Δ                   t   r       -       2   ·   Δ                     t   f             otherwise                   (   9   )                                
     The period-normalized fractional phase is then described by Equation 10 as: 
     
       
         φ F   =Δt   frac   /T   v   (10) 
       
     
     In the present implementation, the fractional phase φ F  is not needed. Instead, Δt r  is used to calculate the ε(k) correction of Equation 5 that is positive and ε∈ (0,1). Δt r  has to be normalized by dividing it by the clock period, in order to properly combine it with the integer phase detector output, θ d .                ɛ        (   k   )       =       Δ                       t   r          (   k   )       /       T   v          (   k   )           =     {           Δ                     t   r     /   2          (       Δ                   t   f       -     Δ                   t   r         )               Δ                   t   r       ≤     Δ                   t   f                   Δ                     t   r     /   2          (       Δ                   t   r       -     Δ                   t   f         )           otherwise                     (   11   )                                
     When the dVCO  104  clock period T v  is an integer division of the frequency reference clock period T r , the ε(k) samples are seen to be constant. The ε(k) samples increase linearly within the modulo (0,1) range where this ratio is fractional. In view of the foregoing, a simple pattern can therefore be easily predicted in digital form that closely corresponds mathematically to the well-known analog fractional phase compensation scheme of fractional-N PLL frequency synthesizers. 
     
       
         {tilde over (ε)}( k )=ε( k )−fract(θ r ( k ))  (12) 
       
     
     The composite phase error θ e (k) is obtained through correcting the integer-valued θ d (k) by fractional-division-ratio-corrected ε(k) as shown in Equation 13. 
     
       
         θ e ( k )=θ d ( k )−{tilde over (ε)}( k )  (13) 
       
     
     The fractional phase detector output ε(k) or φ F (k) sequence can be easily compared on a bit-by-bit basis; and since the expected output pattern is known in advance and is now in the digital format, a better alternative of a Viterbi sequence detection or a matched filter could be used. In such a scenario, the space difference between the observed and expected patterns could be output as the fractional phase error. This solution provides a system with less reference feed through and lower overall error. 
     The present PLL loop operation can be further enhanced by taking advantage of the predictive capabilities of the all-digital PLL loop. The dVCO  104 , for example, does not necessarily have to follow the modulation FCW  116  command with the normal PLL loop response. In one embodiment, where the dVCO  104  control and the resulting phase error measurement are in numerical format, it is easy to predict the current K vco  gain of the dVCO  104  by simply observing the past phase error responses to the NCO corrections. With a good estimate of the K vco  gain, the normal NCO control could be augmented with the “open loop” instantaneous frequency jump estimate of the new FCW  116  command. It can be appreciated that the resulting phase error should be very small and subject to the normal closed PLL loop correction transients. 
     Since the time response of this “type 1” PLL is very fast (less than 1 μsec), the prediction feature is less important for channel hopping, where the allowed time is much greater. The foregoing prediction feature is, however, essential to realize the direct frequency synthesizer modulation in the  Gaussian frequency shift keying  GFSK modulation scheme of “BLUETOOTH” or GSM. 
     In view of the above, it can be seen the present invention presents a significant advancement in the art of RF synthesizer circuits and associated methods. This invention has been described in considerable detail in order to provide those skilled in the RF synthesizer art with the information need to apply the novel principles and to construct and use such specialized components as are required. In view of the foregoing descriptions, it should be apparent that the present invention represents a significant departure from the prior art in construction and operation. However, while particular embodiments of the present invention have been described herein in detail, it is to be understood that various alterations, modifications and substitutions can be made therein without departing in any way from the spirit and scope of the present invention, as defined in the claims which follow. For example, while certain embodiments set forth herein illustrate various hardware implementations, the present invention shall be understood to also parallel structures and methods using software implementations as set forth in the claims.