Abstract:
A self-oscillating DCM is disclosed comprising two inductors that charge and discharge 180 degrees out of phase such that the charging inductor is conducting an upward ramping current and the discharging inductor is conducting a downward ramping current. A load receives the upward and downward ramping currents, which combine to create a constant current. The current source that powers the DCM is current limited so as to output a maximum direct current of I IN . A relatively small capacitor is connected across the input terminals of the DCM and allows the inductors to ramp up to a peak current of 2*I IN . Since the current source only supplies the ramping current to one of the inductors at a time up to 2*I IN , and the average current conducted by each inductor is I IN , the current supplied by the current source is a constant I IN . However, since the inductors simultaneously supply two oppositely ramping currents to the load, the load current is a constant current equal to 2*I IN . So the DCM doubles the supply current and halves the average input voltage.

Description:
FIELD OF INVENTION 
   This invention relates to current converter technology and, in particular, to a direct current multiplier (DCM) that generates very low noise. 
   BACKGROUND 
   Various techniques are known to multiply a current. A current multiplier generates a current at an output that is higher than the current supplied by the power supply. Since power is conserved, an increased current output results in an output voltage that is lower than the input voltage. 
   Some current multipliers use a pulsed switching technique controlled by an oscillator, where large filters are used to convert a pulsed waveform into a DC current. Such pulsed multipliers generate electromagnetic interference (EMI) and other electrical noise and are relatively large. One such multiplier is a switch-mode DC-DC converter. In some situations, a substantially noiseless converter is required, precluding the use of a switch-mode converter. 
   A CUK converter is a special topology of DC/DC converter which uses inductive and capacitive energy transfer to generate an output current, with the advantage of low ripple current at the input when supplying a constant load current at the output. For a simple CUK converter, two inductors, a large coupling capacitor, an output filter capacitor, a switch transistor, a diode, and its control circuit are needed. The coupling capacitor must be relatively large for medium and high load currents. 
   In one example of the need for a current multiplier, a high brightness light emitting diode (LED) only needs a small voltage (e.g., 3.4 volts) but a fairly high current. If the power source is a 12 volt car battery, a converter is used to supply the required current through the LED at 3.4 volts. An idealized converter will thus multiply the current drawn from the battery by 12/3.4. However, known converters generate noise (e.g., a switching converter) or require large coupling capacitors (e.g., CUK converter). 
   What is needed is a direct current multiplier that is simple, small, and does not generate substantial noise to both the power supply and the load. 
   SUMMARY 
   A self-oscillating DCM is disclosed comprising two inductors that charge and discharge 180 degrees out of phase such that the charging inductor is conducting an upward ramping current and the discharging inductor is conducting a downward ramping current. A switching circuit alternately charges and discharges the inductors. A small capacitor is connected across the DCM input terminals. A load receives the upward and downward ramping currents, which combine to create a constant current. A filter capacitor is not needed to filter the inductor waveforms into the load. 
   The current source that powers the DCM is current limited so as to output a constant maximum current of I IN . By using a preferred capacitor value across the DCM input terminals, the peak current conducted by each of the inductors is 2*I IN . The average current through each inductor is I IN . The capacitor charges when the inductor ramping current is below I IN  and discharges into the inductor when the inductor ramping current is greater than I IN . Since the current source only supplies the ramping current to one of the inductors at a time, while the capacitor is charging and discharging to make up the difference between the ramping current and the supply current, the current supplied by the current source is a constant I IN . Since the inductors simultaneously supply two oppositely ramping currents to the load, while one inductor is charging through the load and the other inductor is discharging through the load, the load current is a constant current substantially equal to 2*I IN . So the DCM effectively doubles the supply current and halves the average voltage from the input to the load. 
   DCMs may be cascaded to further multiply the current. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a block diagram of a DCM in accordance with the invention. 
       FIG. 2  is a schematic diagram of one embodiment of the DCM of  FIG. 1 , which doubles the input current. 
       FIG. 3  is a flowchart identifying various steps performed by the circuit of  FIG. 2  in its steady state. 
       FIGS. 4A-4D  illustrate the current waveforms (current vs. time) at various nodes in the circuit of  FIG. 2 . 
       FIG. 5  is a diagram of the circuit of  FIG. 2  where a current-limited LDO regulator supplies the current to the DCM and synchronous rectifiers are used. 
       FIG. 6  is a circuit similar to  FIG. 2  but using low-side NMOS transistors. 
       FIG. 7  illustrates how multiple DCMs can be cascaded to provide additional current multiplication. 
   

   DETAILED DESCRIPTION 
     FIG. 1  illustrates one embodiment of the invention where the direct current multiplier  10  doubles the current from the current source  12 . The current to the load  14  is therefore I L =2*I IN , and the output voltage is V L =V IN /2. The DCM  10  can also be configured to multiply the current by other factors. The current source  12  can be any current limited supply that supplies a constant current I IN . The current source  12  may be a fixed current source or a current limited voltage source (both being referred to herein as a current source). One way to supply a current-limited current is to use a transistor whose current is sensed and limited to the intended maximum current. 
     FIG. 2  illustrates one embodiment of the DCM  10 , which is a self-resonating circuit. The load in the example is a high brightness light emitting diode (LED)  16  that drops about 3.4 volts with a current through it of 350 mA. The current source supplies I IN =175 mA. The operation of the DCM  10  of  FIG. 2  will be described with reference to the flowchart of  FIG. 3  and the current waveforms of  FIGS. 4A-4D . 
   FIGS.  3  and  4 A- 4 D assume a capacitor C 1  value of 330 nF and inductor values of 100 μH. The capacitor value is high enough to receive current from the current source and supply current to the inductors to enable the inductors to reach the full peak current of 2*I IN  per inductor. With significantly smaller values of the capacitor, current to the load will still be doubled, but peak current will decrease and ripple will increase along with other parameters changing. 
   A rule of thumb to determine a preferred capacitor C 1  value may use the following relationship: C 1 ≦L*(I L ) 2 /(V L ) 2 , where L=L 1 =L 2 . 
   If the capacitor value is much larger than the rule of thumb value, the peak inductor current could be more than double the input current. 
   A current source supplies an input current I IN  to node  18  at a voltage of V IN  (step  19 ). 
   It will be assumed that the cross-coupled PMOS transistors Q 1  and Q 2  are in a state where Q 1  is on and Q 2  is off. In step  20  of  FIG. 3 , the inductor L 1  charges through the series connection of transistor Q 1 , inductor L 1 , and LED  16 . This results in a ramping current I 1  through inductor L 1  ( FIG. 4A ) from time T 0  to T 1 . Since the transistor Q 1  node  22  is held at a high voltage (almost V IN ) during this time, and node  22  is connected to the gate of transistor Q 2 , transistor Q 2  is off. 
   Prior to inductor L 1  being charged, it is assumed inductor L 2  had been charged from the previous switching cycle until its current I 2  reached 2*I IN . While transistor Q 1  is on and inductor L 1  is being charged, transistor Q 2  is off, and inductor L 2  is discharging though the circuit formed by LED  16  and the forward biased diode D 2  (step  24 ). The diodes D 1  and D 2  are preferably Schottky diodes to achieve a low forward drop. Alternatively synchronous rectifiers may be used to achieve virtually zero voltage drop. As inductor L 2  is discharging, the voltage at node  25  is one diode drop below ground. The inductor L 2  discharges at the same rate at which it was charged, so it generates a ramping down current, as shown in  FIG. 4B  from time T 0  to T 1 , at the same time that the inductor L 1  is conducting a ramping up current ( FIG. 4A ). The inductance values of inductors L 1  and L 2  are the same. 
   The capacitor C 1  stores charge (indicated as a positive current I C  in  FIG. 4C ) from the current source while the ramping up current is less than I IN  and discharges into the inductor (indicated as a negative current I C  in  FIG. 4C ) when the ramping up current exceeds I IN . 
   As shown in step  26 , the total current supplied through the LED  16  from inductors L 1  and L 2  is a steady 2*I IN . The combined inductor waveforms are shown in  FIG. 4D  as a steady current through the LED  16  equal to 2*I IN . 
   If the inductor values are not equal, they will behave differently. The basic function of the DCM will be the same, but the current transmission ratio I L /I IN  will be less than 2. For adjusting the output current, it is better to adjust the input current rather than using unmatched inductor values. 
   When the ramping current I 1  reaches its possible maximum (in this case 2*I IN ), the ramping will stop, and the voltage at node  22  will drop to approximately the voltage at node  28  (due to v=Ldi/dt). It is assumed in the example that the LED  16  is dropping 3.4 volts, so the voltage at node  28  is 3.4 volts. Since the gate voltage of transistor Q 2  is now well below its source voltage of 6.8 volts (because the DCM is a current doubler), transistor Q 2  will turn on, which raises the voltage at node  25  to approximately 6.8 volts to turn off transistor Q 1  (step  32 ). 
   Now that transistors Q 1  and Q 2  have changed states, inductor L 1  discharges (step  34 ) through the LED  16  and diode D 1  is forward biased, and inductor L 2  charges (step  36 ) due to the series connection of transistor Q 2 , inductor L 2 , and LED  16 . Since diode D 1  is now forward biased, the voltage at node  22  is pulled down to one diode drop below ground. The inductor waveforms are shown in  FIGS. 4A-4C , and the combined waveforms are shown in  FIG. 4D  as a steady current through the LED  16  equal to 2*I IN  (step  38 ). 
   As soon as inductor L 2  conducts 2*I IN , the voltage at node  25  goes low to switch transistors Q 1  and Q 2 , and the self-resonating process repeats (step  40 ). For an optimized value of the capacitor C 1 , the time for one half cycle can be calculated (idealized) as T H =L*ΔI 1 /V L , where L is the inductance of either matched inductor, I L  is the load current, ΔI 1  is the inductor current change (equal to 2*I IN ), and V L  is the voltage across the load (equal to the driving voltage across the inductors). Therefore, the oscillator frequency is f=1/(2*T H )=V L /(2*L*ΔI 1 )=V L /(4*L*I IN ). For the component values mentioned above, the oscillating frequency can be estimated to be about 40-50 kHz. 
   Since only one inductor is being charged at a time, the current drawn from the current source  12  ( FIG. 1 ) is I IN . However, since the current through the LED  16  is a combination of the two inductors L 1  and L 2  contributing increasing and decreasing currents (I 1  and I 2 ), offset by 180 degrees, the current through the LED is 2*I IN , which is double the current from the current source  12  ( FIG. 4D ). 
   Since the current is doubled, and the voltage across the LED  16  is 3.4 volts at the double current, the average voltage at the input node  18  is about 6.8 volts. Because of the charging and discharging of the capacitor C 1 , the voltage at the input node  18  will contain some ripple, where the ripple magnitude is dependent on the component values used in the DCM. 
   The small capacitor C 1  connected across the DCM input terminals helps start the oscillator, stabilizes the oscillator, prevents switching noise being reflected back to the current source, affects settling time, affects peak inductor current, affects efficiency, and affects ripple of the load current. The value of capacitor C 1  may be optimized for a desired inductor peak current (e.g., 2*I IN ). As a result, the LED current and the input current to the DCM have very low ripple. Any ripple can be further reduced by providing a capacitor across the load. 
   The current source  12  ( FIG. 1 ) may take many forms such as a battery or a DC/DC converter that includes current limiting circuitry.  FIG. 5  illustrates the current-limited current source being an LDO regulator  42  (also called a linear regulator). The LDO regulator  42  contains a series transistor  44 , which may be either a MOSFET (in the simplest case a depletion mode MOSFET) or a bipolar transistor. A power supply, such as a car battery, supplies a voltage and current (Ips) to an input port of the LDO regulator  42 . The series transistor  44  is connected between the input port and the output port, supplying all the current to the DCM. The transistor  44  is designed to limit at the intended current so that no further control voltage to the gate or base of transistor  44  increases the current beyond I IN . 
     FIG. 5  shows a simple current limit feedback circuit. The current I IN  passes through a low value resistor R. A first differential amplifier  50  senses the voltage drop across the resistor R, which is proportional to the current. Any other technique for detecting the current can be used instead. A second differential amplifier  52  senses the difference between the actual current and a reference signal  54 . The output of the differential amplifier  52  controls the transistor so that the current feedback signal matches the reference signal  54 . 
   The LDO regulator  42  output voltage will have a ripple, with the average voltage being about twice the load voltage. 
   LDO regulators are less efficient than switch-mode converters at medium to high current levels due to the voltage drop across the series transistor (power loss=voltage drop multiplied by current). Since the power wasted by an LDO regulator is directly proportional to the current conducted by the series transistor, reducing the current through the series transistor by one-half doubles the efficiency of the circuit of  FIG. 5 . 
   The LDO regulator  42  and DCM  10  may be integrated on a single chip. However, inductors will usually be provided external to the chip. The inductors are not magnetically coupled. If Schottky diodes D 1  and D 2  were augmented with synchronous rectifiers (switched transistors with zero voltage drop), the DCM would have virtually 100% efficiency and a current transfer ratio I L /I IN  of 2. The synchronous rectifiers may be NMOS transistors Q 3  and Q 4  whose gates are coupled to the respective gates of PMOS transistors Q 1  and Q 2  such that, when a high signal applied to the gate of transistor Q 1  turns transistor Q 1  off, the high signal applied to the NMOS transistor Q 3  switches transistor Q 3  on. Conversely, when a high signal applied to the gate of transistor Q 2  turns transistor Q 2  off, the high signal applied to the NMOS transistor Q 4  switches transistor Q 4  on. The transistors and couplings would be designed to prevent shoot-through current. The oppositely switching transistor pairs Q 1 /Q 3  and Q 2 /Q 4  in  FIG. 5  may each be a CMOS power inverter. Good transistor matching may allow elimination of the Schottky diodes. 
   A synchronous rectifier may instead be formed as a transistor which is switched on or off based on a polarity of the voltage across its terminals. 
   Circuit simulations confirmed that the efficiency of the DCM can be as high as 99.8%, providing an actual current transfer ratio of about 1.99. 
   The current-limited current supply  42  in  FIG. 5  may even be a switch-mode power supply, where the current supplied is one-half of the current driving the load. This increases the efficiency of the power supply, lowers ripple, and lowers noise. The DCM and switch-mode power supply can be integrated on the same chip. 
   The DCM can even be driven by solar cells or other current generators. 
   Instead of the load being an LED or other conventional load (resistive or non-linear), the DCM may drive a conventional LDO regulator, which outputs a regulated voltage to a load. The LDO regulator drives the load at twice the input current of the DCM. Such an arrangement allows one to supply the DCM (located in-between) by a non-current-limited voltage source/battery because the load current normally is set by the LDO regulator&#39;s regulated voltage divided by the load resistance (I L =V L /R L ). For stable behavior, because of voltage ripple injected into the capacitor C 1 , the DCM should be decoupled from the battery by, for example, a diode. 
     FIG. 6  illustrates the DCM where the transistors Q 1  and Q 2  are low-side NMOS transistors. When transistor Q 1  is on, inductor L 1  is charged through the series connection of the LED  16 , inductor L 1 , and transistor Q 1 . During discharge, inductor L 1  conducts through the LED  16  and forward biased diode D 1 . When transistor Q 2  is on, inductor L 2  is charged through the series connection of the LED  16 , inductor L 2 , and transistor Q 2 . During discharge, inductor L 2  conducts through the LED  16  and forward biased diode D 2 . The operation is the same as described in  FIG. 3 . 
   Various other switching arrangements may be used. 
     FIG. 7  illustrates how multiple DCMs (DCMs  1 - 3 ) can be cascaded to provide additional current multiplication. Each DCM doubles the current, and the resulting voltage is halved. Such arrangements are particularly suitable when high voltage power supplies are used for powering high current loads. 
   The voltage/current source for the DCM may also supply a voltage that is negative with respect to ground. It would be understood that, since the direction of current flow would be reversed, the applicable components (e.g., diodes, LED) would have to be connected in opposite directions, and opposite type transistor would be used. The operation of the DCM will be the same. Alternatively, the negative current source may be connected to the ground terminal in the various figures, with the other terminal connected to ground. 
   Having described the invention in detail, those skilled in the art will appreciate that given the present disclosure, modifications may be made to the invention without departing from the spirit and inventive concepts described herein. Therefore, it is not intended that the scope of the invention be limited to the specific embodiments illustrated and described.