Abstract:
A low level secondary communication signal is summed with an existing primary communication signal in a manner that prevents interference to the existing primary communication signal while providing secondary communication signal benefits such as increased data rate, range, or interference immunity. Examples are presented in which a M-QAM secondary signal is summed with either an ATC Mode S PPM reply or DPSK interrogation primary signal. Legacy Mode S transponders, TCAS, and ADS-B equipment continue to demodulate and decode the primary signal information in accordance with preexisting formats while new enhanced equipment obtains the benefits of the secondary signal.

Description:
TECHNICAL FIELD OF THE INVENTION 
     The present invention relates to the field of communications. More specifically, the present invention relates to summing a low level secondary communication signal to an existing primary communication signal in a manner that prevents interference to the existing primary communication signal while providing an overall communications benefit such as increased data rate, range, or interference immunity. In particular, this invention is applied to air traffic control (ATC) related communications. 
     BACKGROUND OF THE INVENTION 
     Most communication systems in widespread use today evolved from standards and equipment developed many years ago. This is especially true in the field of aircraft communications where equipment fielded one or two decades ago is still in use. Although these systems could be greatly enhanced using today&#39;s advances in communication technology, the cost of simply replacing them would be exorbitant due to the large quantity of this legacy equipment still in use. 
     However, newly manufactured equipment can be enhanced using advances in communication technology if these enhancements can be implemented in a manner that does not degrade the operation of current legacy equipment. It is acceptable that legacy equipment not be able to use or benefit from these enhancements as long as the enhancements do not degrade the performance of the legacy equipment. 
     One such ATC legacy system is the Mode S transponder whose reply data format is implemented using pulse position modulation (PPM). PPM is a modulation technique in which each bit interval is divided into two sub-intervals. A pulse is transmitted in one of the sub-intervals but not in both. Transmitting the pulse in the first sub-interval represents a “1” bit and represents a “0” bit when transmitted in the second subinterval. 
     PPM was initially chosen because it is very simple to both modulate and demodulate. It is a non-coherent modulation technique which ignores the phase of the modulation pulse. An opportunity therefore exists to enhance new Mode S equipment by phase modulating the PPM pulses. 
     Such a PPM pulse phase modulation technique is described in U.S. Pat. No. 8,031,105 “Systems and Methods for Enhanced ATC Overlay Modulation”. This patent describes an empirical based approach of using a second modulator, following the current PPM modulator, to phase modulate individual amplitude modulated output pulses of the first modulator with phase information. 
     Although this technique enhances the current Mode S transponder reply signal by converting the existing amplitude modulated pulses to Phase Shift Keying (PSK) modulated pulses, the technique is limited by the Mode S spectrum and pulse shape requirements. Careful adherence to pulse rise times and PSK modulation rate must be maintained to avoid violating Mode S signal specifications. Many of these phase modulation tradeoffs require empirical simulation and testing to ensure compliance which limits the benefits obtainable using this PPM remodulation technique. 
     Accordingly, it is the object of the present invention to disclose methods and apparatus which provide new and improved techniques for enhancing legacy communication systems without interfering with installed legacy equipment. 
     SUMMARY OF THE INVENTION 
     Briefly, to achieve the desired object of the present invention, a low level communication signal (secondary signal) is summed to an existing unaltered communication signal (primary signal) in a manner that prevents interference to the primary signal while providing an overall communication benefit such as increased data rate, range, or interference immunity. 
     In particular, a low level M-ary Quadrature Amplitude Modulation (M-QAM) secondary signal is summed with an ATC existing unaltered primary signal at a power level, frequency spectrum, phase, and time period chosen to prevent any interference to the primary signal by the secondary signal. 
     In the preferred implementation, the secondary signal is incorporated into new enhanced ATC communication equipment. 
     However, the secondary signal could also be implemented in completely separate external hardware suitably interfaced to new enhanced ATC communication equipment if desired. 
     Other objects and advantages of the present invention will become obvious as the preferred embodiments are described and discussed below. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  illustrates a standard prior art 1090 MHz Mode S PPM reply waveform. 
         FIG. 2  shows a constellation diagram for the Mode S reply PPM waveform of  FIG. 1 . 
         FIG. 3  shows a block diagram of a prior art overlay modulation technique. 
         FIG. 4  shows a constellation diagram for a prior art overlay modulation technique. 
         FIG. 5  shows a functional block diagram of a secondary signal generator applied to a Mode S transponder transmitter. 
         FIG. 6  shows a constellation diagram for 4-QAM. 
         FIG. 7  shows the constellation diagram when a 4-QAM secondary signal is summed with a PPM primary signal. 
         FIG. 8  illustrates the combined signal power shift when the primary and secondary signals are summed. 
         FIG. 9  illustrates the combined signal phase shift when the primary and secondary signals are summed. 
         FIG. 10  illustrates the equivalent secondary signal received power versus range in NM for various secondary signal modulations. 
         FIGS. 11   a  and  11   b  show versions of 16-QAM and 64-QAM phase point constellations respectively for M-QAM pseudo-orthogonal QPSK modulation. 
         FIG. 12  illustrates a standard prior art 1030 MHz Mode S DPSK interrogation waveform. 
         FIG. 13  shows a constellation diagram for the Mode S DPSK interrogation waveform. 
         FIG. 14  shows the constellation diagram when a 4-QAM secondary signal is summed with a DPSK primary signal. 
         FIG. 15  shows a functional block diagram of a secondary signal decoder applied to a Mode S transponder receiver. 
         FIG. 16  shows a secondary signal message format prior to M-QAM encoding, after encoding (transmitted message packet), prior to decoding, and after decoding. 
         FIG. 17  shows an editing ring of radius “r” used to discard secondary signal received symbols that fall outside the editing ring. 
         FIG. 18  illustrates the creation and termination of M-QAM pseudo-orthogonal QPSK modulation decoder paths. 
         FIG. 19  shows a functional block diagram of the secondary communication signal implemented in external hardware. 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Mode S transponder PPM replies on 1090 MHz are defined in RTCA/DO-181C specification “Minimum Operational Performance Standards for Air Traffic Control Radar Beacon System/Mode Select (ATCRBS/Mode S) Airborne Equipment” which is incorporated by reference herein in its entirety. 
     A Mode S reply waveform  100  corresponding to bit sequence (0010 . . . 001)  105  from this specification is illustrated in  FIG. 1 . This waveform consists of a preamble  110  followed by data block  115  which consists of either 56 or 112 PPM data bit intervals  120 . A constellation diagram  150  is illustrated in  FIG. 2  for this modulation signal. Constellation point  155  at zero amplitude and phase indicates no pulse is transmitted while constellation point  160  indicates an amplitude modulated pulse at zero phase is transmitted during preamble period  110  or data block period  115 . 
     A block diagram of a prior art enhancement technique using overlay modulation, which was described in U.S. Pat. No. 8,031,105, is illustrated in  FIG. 3 . Carrier signal generator  205  creates unmodulated carrier  210  which is PPM modulated in PPM modulator  215  by PPM modulator signal  220  from PPM modulator signal generator  222  to produce PPM modulated signal  225  that complies with existing standards. PPM Modulated signal  225  which has previously been modulated is then further modulated in overlay modulator  230  by overlay modulation signal  235  from overlay modulator signal generator  237  to produce modulated ATC stream  240  that provides a reply/squitter signal that is compatible with existing hardware yet contains information in excess of that defined by current transponder standards. ATC stream  240  is transmitted using Mode S transmitter  245 . 
     A constellation diagram  250  of modulated ATC stream  240  is illustrated in  FIG. 4 . Constellation point  155  at zero amplitude and phase indicates no pulse was transmitted. Constellation points  160  and  255  indicate an amplitude modulated pulse at either zero phase or 180 degree phase respectively is transmitted during preamble period  110  or data block period  115  in  FIG. 1  when overlay modulation signal  235  is PSK. That is, amplitude modulated PPM pulses  225  are converted to PSK pulses  240  by overlay modulation  235  as was illustrated in  FIG. 3 . 
     The current invention uses an enhancement method entirely different from this prior art enhancement method. A block diagram  300  of the current invention as applied to the same Mode S transmitter application (that was illustrated in  FIG. 3 ) is illustrated in  FIG. 5 . Carrier signal generator  305  creates unmodulated carrier  310  which is PPM modulated in PPM modulator  315  by PPM modulator signal  320  from PPM modulator signal generator  322  to produce PPM modulated primary signal  325  that complies with existing standards. 
     Carrier reference signal  335  from carrier signal generator  305  is used by secondary signal generator  345  to create low level secondary signal  350 . Power summer  330  combines primary signal  325  with secondary signal  350  to create combined signal  355  which is transmitted using Mode S transmitter  360 . Reference signal  335  and timing signal  340  are used by secondary signal generator  345  to create secondary signal  350  at the proper frequency, phase, and timing. 
     A constellation diagram  375  of a standard 4-QAM (QPSK) secondary signal with amplitude-phase points  380  is illustrated in  FIG. 6 . When this secondary signal  350  is summed with standard PPM signal  325  whose constellation was illustrated in  FIG. 2 , constellation diagram  400  illustrated in  FIG. 7  is created. Constellation points  380  indicate the vector sums of constellation point  160  in  FIG. 2  with constellation points  380  in  FIG. 6 . Note that this summation simply transfers the center of the secondary signal  350  constellation illustrated in  FIG. 6  to the primary signal  325  amplitude indicated by constellation point  160  in  FIG. 2 . 
     The primary benefits of the current invention are that any M-QAM or other modulation can be used for secondary signal  350  and many commercially available standard M-QAM modulation and demodulation integrated circuits can potentially be used if desired. Also, the spectrum and performance of M-QAM modulations are well known by those skilled in the art. Since signal summation is a linear process, the existing Mode S PPM signal spectrum and the secondary signal spectrum simply combine without generating any new signal components as occurs with modulators. Compliance with legacy specifications is straight forward by simply lowering the amplitude of secondary signal  350  as needed to meet spectrum and other specification requirements. 
     Summing a low level M-QAM secondary signal with a Mode S PPM pulse primary signal alters the amplitude and phase of the PPM pulses.  FIG. 8  and  FIG. 9  illustrate the magnitude of the amplitude and phase variations respectively of a primary signal versus the power level of a summed secondary signal. 
     In  FIG. 8 , curves  405  and  410  indicate the increase and decrease of combined signal  355  (in dB along the y-axis) when an in-phase and out-of-phase respectively secondary signal  350  (whose power level in dB below primary signal  325  is indicated along the x-axis) is summed with primary signal  325 . 
     In  FIG. 9 , curve  415  indicates the phase shift of combined signal  355  (in degrees along the y-axis) when a quadrature secondary signal  350  (whose power level in dB below primary signal  325  is indicated along the x-axis) is summed with primary signal  325 . 
     The Mode S spec indicates that the PPM pulse amplitude variation over the duration of the message be no greater than 2 dB. As a M-QAM secondary signal  350  constellation point may either be in-phase or out-of-phase with primary signal  325 ,  FIG. 8  indicates that secondary signal  350  must be at least 7 dB below primary signal  325  to meet the 2 dB specification. 
     M-QAM refers to m-ary Quadrature Amplitude Modulation where m equal 2 is Binary Phase Shift Keying (BPSK), m equal 4 is Quadrature Phase Shift Keying (QPSK), m equal 16 is 16-QAM, and m equal 64 is 64-QAM. If 4-QAM is used for secondary signal  350 , a Signal-to-Noise ratio (SNR) of approximately 8 dB is required for a symbol error rate of 10 −4 . The probability of making at least 1 symbol error per a 112 symbol PPM message at this symbol error rate is 0.011 which indicates a loss of approximately 1 out of every one hundred 112-symbol messages. Mode S PPM data implements error detection but not error correction. A single symbol error causes a message error and the Mode S specifications require a 90 percent reception probability. Requiring a symbol error rate of 10 −4  provides adequate implementation margin to ensure the 90 percent reception requirement is met. 
     In contrast, a SNR of approximately 16.5 dB is required for a legacy PPM non-coherent symbol error rate of 10 −4 . If Mode S used 4-QAM instead of non-coherent PPM, it would be 8.5 dB (16.5 dB−8 dB) more sensitive. Therefore, if 4-QAM secondary signal  350  at a power level 8.5 dB below legacy PPM primary signal  325  is summed with primary signal  325 , the secondary signal  350  symbol and message error rate will be identical to legacy PPM primary signal  325  symbol and message error rate. However, secondary signal  350  carries 2 bits per symbol compared to 1 bit per symbol for legacy PPM primary signal  325  so the data rate of this 4-QAM secondary signal communications is twice that of the current legacy Mode S PPM primary signal communications. 
     Using 16-QAM requires a SNR of 13 dB for a symbol error rate of 10 −4 . Selecting 16-QAM for secondary signal  350  would increase the secondary signal communications rate to 4 times the Mode S PPM rate but its sensitivity would be 5 dB lower since 16-QAM requires 5 more dB (13 dB−8 dB) signal power than 4-QAM. Using 16-QAM is equivalent to decreasing received signal power by 5 dB when determining maximum range performance. 
       FIG. 10  illustrates the range penalty due to increased or decreased sensitivity. The Mode S specifications are based on obtaining a 90 nautical mile (NM) reception range when the received signal power level is −84 dBm which is indicated by curve  420 . That is, the required Mode S message error rate can be achieved when the received signal power level is −84 dBm or greater. As is well known, received signal level decreases with range at 6 dB per octave. Curve  425  assumes the Mode S equipment just meets the −84 dBm specification (y-axis) at 90 NM range (x-axis). Curve  430  illustrates range when received signal power is decreased by 5 dB which is equivalent to a 5 dB decrease in sensitivity. Note that maximum range (−84 dBm point) drops from 90 NM to 50 NM. Selecting 16-QAM for secondary signal  350  to obtain 4 times the Mode S PPM data rate may be a good tradeoff against shorter range if most high speed communications occur at ranges under 50 NM. 
     Although any M-QAM modulations can be selected for secondary signal  350 , the pseudo-orthogonal QPSK signal encoders and decoders described in commonly owned U.S. Pat. No. 8,098,773 “Communication Method and Apparatus” and U.S. Pat. No. 8,437,431 “Sequential Decoder Fast Incorrect Path Elimination Method and Apparatus for Pseudo-Orthogonal Coding” are most attractive due to their superior error correcting abilities. U.S. Pat. Nos. 8,098,773 and 8,437,431 are incorporated herein by reference. 
     As described in U.S. Pat. No. 8,098,773, two bits of message data are rate 1/2 Viterbi encoded into 16-QAM symbols using the constellation illustrated in  FIG. 11   a . Symbols are decoded using an efficient sequential decoding algorithm in which all paths through the decoding tree are retained until it is certain that a particular path cannot be the correct path. This rate ½ 16-QAM encoder/decoder provides 2 bits per symbol, does not require additional bandwidth, requires less SNR to achieve a given BER, and has far superior error correction ability with respect to any other short message coding techniques. The rate ½ Viterbi 64-QAM encoder/decoder described in U.S. Pat. No. 8,098,773 provides 3 bits per symbol. Its constellation, illustrated in  FIG. 11   b , contains 36 distinct phase point symbols but implements 64-QAM as explained in U.S. Pat. No. 8,098,773. 
     The decrease in BER with increased SNR is extremely steep for pseudo-orthogonal QPSK. Essentially error free performance in Additive White Gaussian Noise (AWGN) occurs at a SNR of 3 dB for 16-QAM and 6 dB for 64-QAM. Using the same analysis for 16-QAM pseudo-orthogonal QPSK as was used earlier for uncoded 4-QAM, an increase of 13.5 dB (16.5 dB−3 dB) is obtained at the same power level and an increase of 5 dB (13.5 dB−8.5 dB) when secondary signal  350  is 8.5 dB below primary signal  325 . Using 16-QAM pseudo-orthogonal QPSK is equivalent to increasing received signal power by 5 dB. Since 64-QAM requires 3 dB more SNR than 16-QAM, using 64-QAM is equivalent to increasing received signal power by 2 dB. 
     Curve  435  in  FIG. 10  illustrates range when received signal power is increased by 5 dB (which is equivalent to a 5 dB increase in sensitivity) by using 16-QAM pseudo-orthogonal QPSK. Note that maximum range (−84 dBm point) increases from 90 NM to 158 NM. Range is only slightly increased over legacy non-coherent PPM using 64-QAM pseudo-orthogonal QPSK. Note that this analysis is for equipment that just meets the minimum Mode S specifications of 90 NM reception range at −84 dBm. Mode S equipment typically exceeds this range. Exceeding this spec by 3 dB or 6 dB increases the 16-QAM pseudo-orthogonal QPSK range from 158 NM to 224 NM or 316 NM respectively. 
     Automatic Dependent Surveillance-Broadcast (ADS-B) squitters are periodic transmissions by Mode S transponders or stand-alone ADS-B equipment that are primarily used by aircraft to report various navigation, intent, and other data comprising the ADS-B information. Long range reception of ADS-B transmission is important for in-trail following communications where aircraft maintain separation from each other on over-water routes. A 16-QAM pseudo-orthogonal QPSK secondary signal provides 2 bits per symbol and could simply repeat the PPM primary signal data as part of its data package. In this way, in-trail communications could be maintained even though the aircraft are out of primary signal ADS-B communication range. 
     A Mode S transponder interrogation signal on 1030 MHz is also defined in RTCA/DO-181C and its waveform  450  is illustrated in  FIG. 12 . This waveform consists of a number of pulses followed by data block  455  which consists of either 56 or 112 Differential Phase Shift Keying (DPSK) data bits. During each data bit chip  460  either an in-phase or out-of-phase signal is transmitted. A chip is an unmodulated interval preceded by possible phase reversals. If preceded by a phase reversal, a chip represents a “1” bit. If preceded by no phase reversal, a chip represents a “0” bit. 
     A constellation diagram  470  is illustrated in  FIG. 13  for this data block  455  DPSK modulation signal. Constellation point  475  indicates an in-phase (zero degrees) signal is transmitted while constellation point  480  indicates an out-of-phase (180 degrees) signal is transmitted. 
     When constellation diagram  375  of standard 4-QAM secondary signal  350  with amplitude-phase points  380  in  FIG. 6  is summed with DPSK primary signal  325  constellation  470  illustrated in  FIG. 13 , constellation diagram  485  illustrated in  FIG. 14  is created. Constellation points  490  indicate the vector sum of constellation point  475  in  FIG. 13  with constellation points  380  in  FIG. 6  and constellation points  495  indicate the vector sum of constellation point  480  in  FIG. 13  with constellation points  380  in  FIG. 6 . Note that this summation simply transfers the center of the secondary signal  350  constellation illustrated in  FIG. 6  to the primary signal  325  amplitude and phase indicated by constellation points  475  and  480  in  FIG. 13 . 
     The Mode S spec indicates that the DPSK amplitude variation between successive phase modulation chips in data block  455  be less than 0.25 dB and the tolerance on the zero and 180 degree phase relationships be within plus and minus 5 degrees. To meet the amplitude requirement,  FIG. 8  indicates secondary signal  350  amplitude must be at least 12.5 dB below primary signal amplitude  325 . To meet the phase requirement,  FIG. 9  indicates secondary signal  350  amplitude must be at least 10.5 dB below primary signal amplitude  325 . 
     A SNR of approximately 9.5 dB is required for a legacy Mode S interrogation DPSK signal symbol error rate of 10 −4 . If 4-QAM is used for secondary signal  350 , a SNR of approximately 8 dB is required for a symbol error of 10 −4 . If 4-QAM was used instead of DPSK, it would be 1.5 dB (9.5 dB−8 dB) more sensitive. Therefore, if 4-QAM secondary signal  350  at a power level 12.5 dB below legacy DPSK primary signal  325  is summed with primary signal  325 , its sensitivity would decrease by 11 dB (12.5 dB−1.5 dB) indicating 4-QAM is not a good choice for secondary signal  350 . 
     If 16-QAM pseudo-orthogonal QPSK, which can operate at a SNR of 2 dB for a symbol error of 10 −4  with adequate hardware resources is used for secondary signal  350 , it would have 6 dB (8 dB−2 dB) more sensitivity than using 4-QAM. This would reduce the decrease in sensitivity to 5 dB (11 dB−6 dB). Using 16-QAM pseudo-orthogonal QPSK is equivalent to decreasing received signal power by 5 dB as explained earlier. As illustrated by curve  430  in  FIG. 10 , the tradeoff for doubling the DPSK data rate (tripling the combined signal data rate) and reducing message error rate is a reduction in range. 
     Implementing secondary signal  350  in new Mode S transponders is fairly straight forward as is illustrated in  FIG. 5 . Carrier signal generator  305  must generate a stable coherent carrier signal with minimal phase shift over the duration of reply transmission  100  illustrated in  FIG. 1 . This is necessary so that secondary signal  350  can be demodulated at the receiver. In legacy Mode S equipment, the phase of each PPM pulse may be random. 
     In most modern communication equipment, PPM modulator signal generator  322  and secondary signal generator  345  are implemented in software. PPM modulator signal  320  and secondary signal  350  are created as baseband digital signals, digitally upconverted to a convenient intermediate frequency, and digital-to-analog converted prior to driving PPM modulator  315  and signal summer  330  respectively. As such, their relative power and phase relation can be precisely controlled. Combined signal  355  is then upconverted to 1090 MHz, amplified, and transmitted. 
     The generation of M-QAM signals is well known by those skilled in the art and VHDL cores for Field Programmable Gate Arrays (FPGA) are readily available. U.S. Pat. No. 8,098,773 describes the implementation of pseudo-orthogonal QPSK signal encoders in detail. 
     A functional block diagram  500  of a Mode S transponder receiver and secondary signal decoder is illustrated in  FIG. 15 . Mode S receiver  505  receives and downconverts the 1090 MHz combined signal to a convenient IF signal  510 . In most modern communication equipment, IF signal  510  is Analog-to-Digital (A/D) converted using A/D converter  515  into In-Phase (I) and Quadrature-Phase (Q) digital signals  520 . PPM decoder  530  digital filters, removes Doppler phase shift, down converts to baseband, and decodes the PPM primary signal data  535  which is sent to central processor  540 . PPM decoder  530  also sends Doppler removed baseband I,Q digital data  525  to secondary signal decoder  545 . Decoded secondary signal data  550  is sent to central processor  540 . 
     Secondary signal decoder  545  in  FIG. 15  may also contain Random Access Memory (RAM)  555  to temporarily store digitized Doppler removed baseband I,Q digital data  525  from PPM decoder  530  along with dedicated M-QAM decoder chip  560 . This allows a much simpler processor  565  to be used to decode secondary signal messages than if the messages had to be processed in realtime. It also allows baseband I,Q digital data  525  to be processed multiple times to compensate or remove any residual Doppler phase shift. For example, an initial pass could remove any overall message rotation caused by Doppler shift prior to actually decoding the message. 
     The decoding of M-QAM signals is well known by those skilled in the art and VHDL cores for Field Programmable Gate Arrays (FPGA) are readily available. U.S. Pat. Nos. 8,098,773 and 8,437,431 describe the implementation of pseudo-orthogonal QPSK signal decoders in detail. 
       FIG. 5  and  FIG. 15 . illustrate enhancing an existing Mode S PPM reply communication system. However, the method can also be applied to many other existing communication systems. The methodology is to select and generate an appropriate secondary signal, sum it with the existing communication system primary signal to create a new combined signal, and transmit the new combined signal instead of the primary signal. The secondary signal must be selected so that the combined signal still conforms to the existing primary signal system specifications. When the combined signal is received, both the primary and secondary signal information is extracted. 
     The Mode S transponder PPM reply on 1090 MHz and the Mode S transponder interrogation signal on 1030 MHz both send data as either a 56 bit or 112 bit data packet.  FIG. 5  and  FIG. 15 . illustrate enhancing the PPM reply primary signal (primary data packet) using a secondary data packet for the secondary signal in which both the primary and secondary data packets contain the same number of symbols and the symbols are aligned in time. This approach has the advantage that both the PPM primary signal and the M-QAM secondary signal have the same symbol rate so the same integrate and dump matched filter can be used to decode both signals. All downconversion, filtering, and Doppler removal resources can be placed in PPM decoder  530  in  FIG. 15 . Baseband I,Q samples  525  are already optimum filtered and processed so that function does not have to be repeated again in secondary signal decoder  545 . 
     In contrast, prior art enhancement described in U.S. Pat. No. 8,031,105 and illustrated in  FIG. 3  postulate using a higher rate overlay modulation then the existing PPM modulation rate whereby multiple phase transitions are encoded in one logical PPM bit sub-interval. This approach degrades sensitivity and Doppler removal because optimum match filtering over the entire PPM symbol period cannot be used when multiple random data bits occur over this period. 
     A beneficial original, encoded, and decoded message format when pseudo-orthogonal QPSK signal encoders and decoders are selected for secondary signal  350  is illustrated in  FIG. 16 . A fixed length message  600  is defined composed of a fixed length message body  610  and a short fixed length postamble  615 . 
     The purpose of using a fixed message length  600  and postamble  615  is to end encoded message  660  in the encoder zero state. Postamble  615  contains a number of “zero” encoder input bits, the number related to the constraint length of the Viterbi encoder, which returns M-QAM pseudo-orthogonal QPSK signal encoder  605  to its zero state. Encoded message  660  is the actual transmitted data symbol packet and consists of encoded message body  625  and encoded postamble  630 . The final symbols in encoded postamble  630  encoder output, are output symbols resulting from encoder postamble  615  input “zero” bits, obtained as the Viterbi encoder returns to its zero state. 
     The convolutional decoder described in U.S. Pat. No. 8,437,431 retains all paths through the decoding tree until it is certain a path is not the correct path. At the end of the message, in a noisy communication environment, the minimum metric path may not be the correct path. 
     When transmitted encoded message  660  (transmitted symbol packet) is received, a number of noiseless locally generated “zero symbols”  635  are appended to received encoded message  660  prior to decoding. Since appended “zero” symbols  635  are noiseless, they quickly eliminate all incorrect paths thus identifying the correct path in decoder  620  and thus the correct decoded message  670 . The decoded postamble  650  of decoded message  670  will be discarded and the original transmitted message in decoded message body  645  will be sent to central processor  540 . 
     M-QAM pseudo-orthogonal QPSK decodes the entire message as a packet as opposed to decoding each individual symbol. This is an important benefit in a high traffic environment because it allows many individual packet symbols to be jammed and lost yet not make a packet error. As long as the correct path is retained, no message error will occur. The message format illustrated in  FIG. 16  ensures the correct path will be retained when symbol jamming occurs late in the message. 
     Eliminating the effects of jamming on Mode S PPM replies when using 16-QAM pseudo-orthogonal QPSK is illustrated in  FIG. 17  which replaces 4-QAM constellation points  380  in  FIG. 7  with the 16-QAM constellation points illustrated in  FIG. 11   a.    
     Assume 16-QAM constellation symbol  710  in  FIG. 17  was transmitted and point  715  was received due to noise when interference was not present. Point  715  will be meaningful processed in the decoder because it is close to an actual constellation point. However, if interference jamming moves transmitted point  710  to point  725 , it is not meaningful to process point  725  in the decoder. 
     Obvious jammed symbols will be eliminated by modifying the decoder algorithm to in effect place an editing ring  720  of some radius “r” around the center of the constellation and not processing any symbols that fall outside the ring. Current paths in the decoder will be propagated as usual but their current residuals will not be changed. Since a jammed symbol adds no information to the decoding process, eliminating the symbol will not falsely affect the current state of the decoder. Radius “r” can either be fixed or adjusted based on current SNR. The editing ring  720  should be just outside the expected noise altered constellation points. 
     Optimum M-QAM pseudo-orthogonal QPSK decoding in a noisy and high interference environment can be obtained by using editing ring  720  in  FIG. 17 , RAM  555  in  FIG. 15 , and message format  600  in  FIG. 16  as illustrated by decoding path history  800  in  FIG. 18 . Points  825  indicate symbol times along the correct path  820 . All other paths  830  are incorrect paths. As explained in U.S. Pat. Nos. 8,098,773 and 8,437,431, at each symbol time, the residuals of all paths are updated and any existing path spawns new paths. A path exists until its accumulated residual exceeds a threshold, at which time the path is eliminated. 
     As residuals are accumulated, editing ring  720  does not allow the current residual of any path to be corrupted by a jammed symbol. Each current path generates multiple new paths at each symbol time  825 . RAM  555  saves processing resources by allowing the decoder to serially decode a path until it is eliminated instead of requiring enough resources to decode all current paths in parallel. For example path  810  exists for 4 symbol times before it is eliminated. When a path is eliminated, its entire path history can be deleted allowing its processor and memory resources to be used to process another path. 
     Symbol  835  is the last transmitted symbol. Using message format  600  allows noiseless non-jammed locally generated zero symbols  635  to be entered into decoder  620  to flush out all remaining current paths except correct path  820 . The correct path residual in decoder  620  does not change as zero symbols  635  are entered while the path residuals of all incorrect paths increase. Zero symbols  635  can be entered until only one path remains which will be correct path  820 .  FIG. 18  assumes this occurs at symbol  840  where all incorrect paths are eliminated except correct path  820 . 
     For some enhanced communication applications, it may be desirable to implement the secondary communication signal completely separate from the existing communication system as illustrated by functional block diagram  900  in  FIG. 19 . 
     Normally, an existing primary signal communication system (transceiver)  905  is simply connected to antenna  925  through a RF antenna cable. To implement secondary communication, multiport coupler  915  is inserted between transceiver  905  and antenna  925 . Transceiver  905  is connected to transmit port  930  and antenna  925  is connected to receive port  935 . These ports implement a low loss multiport coupler  915  connection between transceiver  905  and antenna  925 . 
     Phase lock loop (PLL) and voltage controlled oscillator (VCO) circuit  955  obtain a sample of transceiver  905  primary signal via transmit output coupling port  940  and generates a carrier reference signal  965  for secondary signal generator  975 . Likewise, demodulator circuit  960  generates a timing signal  970 . These signals are used to create a secondary signal constellation from transmit data  980  that is aligned in time and phase with transceiver  905  primary signal. Receive input coupling port  945  sums the primary and secondary signals to create a combined signal which is transmitted on antenna  925 . A combined signal received on antenna  925  is coupled through receive output coupling port  950  to secondary signal receiver and decoder  985  to extract secondary signal data  990 . 
       FIG. 19  further illustrates that this secondary communication signal invention does not modulate the existing primary communication signal. Instead, the primary and secondary communication signals are simply vector summed. For most applications the preferred implementation is to incorporate the secondary communication signal into an updated version of existing equipment. However, if that is not possible, this enhancement can be implemented completely external of existing equipment. 
     In summary, a key advantage of this secondary communication signal invention over prior art is its ability to implement standard communication signals, such as M-QAM modulation, which have well known performance and spectral characteristics. FPGA code for multiple versions of secondary signal M-QAM modulations could be implemented, stored in FPGA configuration memory, and loaded into FPGA hardware for different communication scenarios. For example, 16-QAM pseudo-orthogonal QPSK could be loaded for long range in-trail communication when over oceans and uncoded 16-QAM or even 64-QAM could be loaded for short range high data rate communications. 
     For data exchange between specific addressed users, the equipment could automatically negotiate with each other to select the highest data transfer rate as do typical modems. Since M-QAM spectrums are primarily a function of symbol rate, higher data rates can be selected simply by loading higher order M-QAM modulation types (higher M number). Transmitting, receiving, downconverting, and match filtering functions would not change since the symbol rate does not change as M-QAM modulation order increases. 
     Although the examples presented herein were for ATC Mode S transponder applications, this invention is applicable for numerous other applications in which adding a secondary communication signal to a legacy primary signal provides a benefit. Many simple modifications to the described system are possible without departing from the spirit of the invention.