Abstract:
The invention concerns a control circuit ( 20 ) of a switch ( 6 ) chopping a voltage supply of a primary winding of a power converter transformer, comprising means ( 45 ) for detecting the current in the switch in closed state after a predetermined time following each closure of said switch, and a comparator ( 40 ) of said current relative to a threshold (Ilim), the result of said comparator being taken into account for a predetermined time interval close to the beginning of a closing cycle of said chopping switch.

Description:
BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The present invention relates to the field of power converters of low voltage switched-mode power supply type. The present invention more specifically applies to isolated power supplies, that is, supplies that have no common point between the input voltage (for example, the A.C. supply system) and the regulated D.C. output voltage. The isolation is obtained by means of a transformer having a primary winding associated with a pulse-width modulation controlled switch, and having a secondary winding associated with a diode and with a capacitor providing the output voltage. 
   2. Discussion of the Related Art 
     FIG. 1  shows a conventional example of a switched-mode power supply of the type to which the present invention applies. Two input terminals P, N receive an A.C. voltage Vac, for example, the mains voltage. Voltage Vac undergoes a rectification, for example a fullwave rectification by means of a diode bridge  1 . The A.C. input terminals of bridge  1  are connected to terminals P and N, and its rectified output terminals  2 ,  3  provide a voltage Vr. Voltage Vr is generally smoothed by means of a capacitor C 1  connected between terminals  2  and  3  which form the input terminals of the actual switched-mode power supply. 
   The converter of  FIG. 1  is a so-called flyback converter in which a transformer  4  with inverted phase points has its primary winding  5  connected in series with a switch  6  between terminals  2  and  3 . The phase point of winding  5  is connected to a terminal of switch  6 , the other terminal of which is connected to terminal  3 . Switch  6  is connected in switched mode and at a non-audible high frequency (generally greater than 20 kHz). A secondary winding  7  of transformer  4  is associated with a capacitor C 2  across the terminals Sp and Sn of which is provided D.C. output voltage Vout. The phase point of winding  7  is connected to terminal Sp by a diode D 1 , the cathode of diode D 1  being connected to terminal Sp. The other terminal of winding  7  is connected to terminal Sn. Ground terminals  3  and Sn are isolated from each other by means of a capacitor Ci. 
   When switch  6  is on, the phase point of winding  7  is at a negative potential. Diode D 1  thus is off and a current is stored in primary winding  5 . Upon turning off of switch  6 , the phase points of windings  5  and  7  both become positive. Diode D 1  is forward biased. Capacitor C 2  is then charged with the power transferred to secondary winding  7 . 
   Switch  6  (for example, a MOS transistor) is, in the example of  FIG. 1 , integrated in a circuit  10  with its electronic control circuit. An example of such an integrated circuit, sold by STMicroelectronics Company, is known under trade name VIPER. Circuit VIPER is comprised of an input terminal Vdd intended for receiving a positive power supply, a voltage reference terminal Vss connected to ground  3 , and a terminal OSC conditioning an oscillation frequency. Circuit  10  further includes a terminal COMP for compensating the regulation loop, connected by a resistor R 5  in series with a capacitor C 5  to ground  3 . Finally, a terminal  12  is connected to the drain of the integrated N-channel transistor, the source of which is connected to terminal Vss. The gate of transistor  6  is connected at the output of a control circuit  11  (CTRL). Circuit  11  includes a comparator (not shown), a first input of which receives an internal voltage reference and a second input of which is connected, internally, to the positive supply terminal. The control, that is, the modification of the width of control pulses of switch  6 , is performed by for example using a loop of regulation of integrated circuit  10 , which attempts maintaining its supply voltage (Vdd-Vss). This type of regulation is performed on the primary side of the transformer. The output voltage may also be regulated based on a measurement on the secondary side, transferred to circuit  10  by a galvanic isolation element (for example, an optocoupler). Terminal OSC is connected to the midpoint of a series association of a resistor R 3  and of a capacitor C 3  between a line  13  of local supply of circuit  10  and the ground. Resistor R 3  and capacitor C 3  set the oscillation frequency. A capacitor C 4  for filtering the local supply voltage is connected between terminal  13  and terminal  3 . 
   A problem which arises in flyback-type converters is that, when a short-circuit occurs at the output of the power converter, diode D 1  and transformer  4  are not protected. They thus risk being damaged under the effect of the strong current that flows through the transformer. Further, a melting of the transformer breaks the galvanic isolation, which is particularly dangerous if the power converter is supplied by the mains. Standards generally determine the short-circuit strength duration of a power converter of this type. 
   In an application to a switched-mode converter, local supply line  13  of circuit  10  is often connected, as illustrated in  FIG. 1 , by a diode D 2 , to the phase point of an auxiliary winding  8  of transformer  4 . In this case, the other terminal of auxiliary winding  8  is connected to reference terminal  3  of the rectified voltage. Auxiliary winding  8  has the function of providing supply voltage Vdd of circuit  10 . The output voltage then is set by the transformation ratio between auxiliary winding  8  and secondary winding  7 . Auxiliary winding  8 , which gives an image of the output voltage, is used, said winding being in direct phase relation with secondary winding  7 . 
   In such an assembly, a current detector (not shown) may be provided in series with switch  6 . The result of the detection is then provided to a comparator which, by means of adapted logic circuits, opens switch  6  when the current exceeds a threshold. The amount of power transmitted to the secondary is thus reduced. Further, since auxiliary winding  8  is in direct phase relation with secondary winding  7 , the voltage drop which appears across the secondary winding is, after a time depending on the value of capacitor C 4 , seen by the supply line of circuit  10 . The supply of circuit  10  becomes insufficient for its operation, which guarantees the turning off of switch  6 . 
     FIGS. 2 ,  3 A,  3 B, and  3 C illustrate the operation of such a power converter in normal state and in short-circuit at the secondary.  FIG. 2  illustrates an example of the shape of voltage VAUX across auxiliary winding  8  in normal operation.  FIGS. 3A ,  3 B, and  3 C respectively show the course of voltage VAUX, of current  1  in switch  6  and of local supply voltage Vdd of circuit  10  when the secondary of the transformer is short-circuited. 
   In normal operation, upon each turning-off (times t 1 ) of switch  6 , voltage VAUX abruptly increases from a negative value to a demagnetization value VDEM. Value VDEM is only reached after a few spurious oscillations associated with the turning-off of switch  6 . Once the demagnetization is complete, voltage VAUX drops (times t 2 ) and exhibits oscillations centered on the voltage zero until the turning-on (times t 3 ) of switch  6  where voltage VAUX becomes negative again. The same waveform is obtained across secondary winding  7 , before the voltage is filtered by capacitor C 2 . 
   When a short-circuit is present between terminals Sp and Sn ( FIGS. 3A to 3C ), the current in switch  6  is limited to a value Imax (times t 10  to t 11 ) by the previously-described detection circuit. Supply voltage Vdd of circuit  10  then progressively decreases ( FIG. 3C ). This progressive decrease reaches a threshold (VddOFF) below which circuit  10  no longer receives the sufficient voltage. From this time t 11  on, switch  6  remains off and no current is transmitted to the secondary. However, this restarts the starting circuit generally associated with the power converter. This restarting causes a progressive increase of local supply voltage Vdd. When this voltage reaches (time t 12 ) the operating threshold (VddON) of circuit  10 , high current surges occur again (times t 12  to t 13 ). Voltage VAUX ( FIG. 3A ) exhibits oscillations for each current peak. 
   The scale is different between  FIG. 2  and  FIGS. 3A to 3C . In  FIG. 2 , the switched mode period (on the order of 10 microseconds) of the supply voltage has been shown. In  FIGS. 3A to 3B , the time interval separating times t 10  and t 11  during which the decrease of local supply voltage Vdd takes several switching cycles and lasts, for example, for approximately 100 milliseconds, has been shown. 
   To respect the standards, a duty ratio that enables the transformers to withstand the high currents Imax ( FIG. 3B ) which, even though they are limited, are much greater than nominal current Inom of normal converter operation. For example, for a nominal current of 2 amperes, the limiting current is on the order of 10 amperes. 
   A problem that remains in this conventional solution is that transformer  4  is still urged at its maximum power. Accordingly, the auxiliary winding generates spurious noise ( FIG. 3A ). This noise is also present in normal state but its amplitude is much greater for a short-circuit of the secondary. Accordingly, although the auxiliary voltage drops, amplitudes sufficient for an autonomous supply of the control circuit are often present. 
   Another problem is linked to the minimum conduction duration of switch  6 . Indeed, it must be possible to turn off switch  6  sufficiently rapidly after each turning-on, otherwise the demagnetization under the low local supply voltage of the auxiliary circuit cannot be performed, and the current is then no longer controlled. 
   SUMMARY OF THE INVENTION 
   The present invention aims at providing a power converter of isolated switched-mode power supply type, which overcomes the disadvantages of known converters and, more specifically, which withstands a short-circuiting of its output terminals. 
   A first solution would be to have the restarting of the starting circuit of the converter switch be conditioned by a complete demagnetization of the auxiliary winding. By detecting the end of the demagnetization, it would be ensured that the converter no longer is in a continuous mode operation (non demagnetized at each cycle). The control of the auxiliary voltage, that is, of the demagnetization point, guarantees the control of the current in the switch. The frequency of circuit  10  can thus be decreased, which results in less current peaks. Further, the problem of switching to a continuous mode is suppressed since the converter is maintained in a discontinuous mode. 
   However, such a solution requires being able to measure the auxiliary voltage. Now, when the switch and its control circuit are integrated, as is the case in a VIPER circuit, no terminal of access to this auxiliary voltage is available. 
   Another disadvantage is that by forbidding an operation in continuous mode, the forming of a bi-voltage converter (220 or 110 volts) is also forbidden. Indeed, in bi-voltage converters, a discontinuous operating mode is generally provided for 220 volts, a continuous operating mode being provided for 110 volts. 
   Thus, the present invention also aims at providing a solution which is compatible with an integration of the control circuit of the switch, more specifically, with an integrated circuit of VIPER type and the corresponding number of terminals. The present invention also aims at providing a solution that enables operation in continuous mode of the converter. 
   More specifically, the present invention provides a circuit for controlling a switch for cutting-up a supply voltage of a primary winding of a power converter transformer, including means for detecting the current in the switch in the on state after a predetermined time that follows each turning-on of said switch, and a comparator of this current with respect to a threshold, the result of said comparator being taken into account for a predetermined duration in the vicinity of the beginning of an on cycle of said switch. 
   According to an embodiment of the present invention, said result is taken into account with a predetermined delay with respect to the turn-on time of the switch. 
   According to an embodiment of the present invention, the circuit includes means for modifying a switching frequency of the supply voltage according to the result provided by said comparator. 
   According to an embodiment of the present invention, the circuit is integrated with the switch. 
   According to an embodiment of the present invention, said current threshold is chosen to allow or not the operation of the converter in continuous mode. 
   The present invention also provides a voltage converter including a circuit for controlling a switch for providing current to a primary winding of a transformer with inverted phase points, a winding of which is associated with a capacitor for providing a regulated D.C. output voltage. 
   According to an embodiment of the present invention, an input terminal of the control circuit intended for setting the oscillation voltage is connected to a supply voltage by a resistor, the value of which is chosen according to a desired minimum switching frequency. 
   The foregoing and other objects, features and advantages of the present invention, will be discussed in detail in the following non-limiting description of specific embodiments in connection with the accompanying drawings, wherein: 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIGS. 1 ,  2 ,  3 A,  3 B, and  3 C, previously described, are intended for showing the state of the art and the problem to solve; 
       FIG. 4  shows, in the form of blocks, an embodiment of an integrated circuit of a power converter according to the present invention; and 
       FIGS. 5A ,  5 B,  5 C,  5 D,  5 E, and  5 F and  6  illustrate, in the form of timing diagrams, the operation of a control circuit according to the present invention. 
   

   DETAILED DESCRIPTION 
   The same elements have been designated with the same references in the different drawings.  FIGS. 2 ,  3 A to  3 C,  5 A to  5 F and  6  are not to scale. For clarity, only those elements of the converter which are necessary to the understanding of the present invention have been shown in the drawings and will be described hereafter. In particular, the internal structure of a VIPER-type integrated circuit has only been detailed for the elements useful to the discussion of the present invention. The rest of its structures, as well as the operation of such a circuit, is known. 
     FIG. 4  shows an embodiment of a control circuit of a supply voltage cut-up switch according to the present invention. The example of  FIG. 4  relates to the case of a circuit  20  integrating switch  6  (for example, an N-channel MOS transistor). This circuit is comprised, for example, of the components integrated in a known VIPER circuit. 
   As previously, and although this is not shown in  FIG. 4 , a circuit according to the present invention belongs to a converter receiving, as an input, a rectified and filtered voltage. This voltage is obtained, for example, by rectifying and filtering an A.C. supply voltage. However, the input voltage may as an alternative come from any supply source, including a D.C. source. 
   Conventionally, switch  6  is connected in series with a primary winding ( 5 ,  FIG. 1 ) of an isolation transformer between the terminals of application of the input voltage. Switch  6  is controlled in pulse-width modulation at a relatively high frequency (for example, several tens of kilohertz). A terminal  12  of integrated circuit  20  corresponds to the drain of transistor  6  and is intended for being connected to the phase point of the primary winding. A terminal  22  of circuit  20  corresponds to a voltage reference terminal Vss, intended for being connected to the reference potential of the input voltage (terminal  3 ,  FIG. 1 ). Circuit  20  is supplied by a low D.C. voltage Vdd. This voltage is, as previously, recovered across an auxiliary winding ( 8 ,  FIG. 1 ) of the transformer. The auxiliary winding is, in the implementation of the present invention, in direct phase relation with a secondary winding of the transformer which provides the regulated D.C. output voltage. The means used to recover voltage Vdd are, for example, formed of capacitor C 4  and of diode D 2  shown in  FIG. 1 . 
   Local supply voltage Vdd is also provided to a terminal  23  of circuit  20 . It further supplies a series association of a resistor R 3  with a capacitor C 3 , the midpoint of which is connected to terminal  24  used to set the oscillation voltage of circuit  20 . 
   Switch  6  is controlled by a signal Vg applied on the gate of the MOS transistor. Signal Vg is provided by a conventional circuit  25  (PWM) intended for modifying the width of the control pulses of transistor  6  to regulate the output voltage of the converter. In practice, circuit  25  includes, among others, a comparator receiving as an input an integrated reference voltage and supply voltage Vdd (or a voltage proportional thereto). The comparator is associated at its output to a current source controlled according to the signal present on a terminal  26  of circuit  20 . Terminal  26  is for example connected, as in  FIG. 1 , to ground  3  by a resistor R 5  in series with a capacitor C 5  which take part in the regulation loop. Terminal  26  corresponds, internally, to the output of an error amplifier of the regulation loop. Circuit  25  further receives an oscillation frequency from a block OSC  27 , the input of which is connected to terminal  24 . The details constitutive of circuit  25  have not been shown and are no object of the present invention. 
   To perform the function of limiting the current in switch  6  in the on state, circuit  20  includes a comparator  30  having a non-inverting input connected to an element  31  for measuring the current in switch  6  and its inverting input connected to terminal  26 . Element  31  measures, for example, the source current of transistor  6  and converts it into a voltage to transmit the information to comparator  30 . The voltage level of the inverting input of comparator  30  corresponds to the output level of the error amplifier of the regulation loop which is, by construction, limited to a maximum voltage. As an alternative, terminal  26  may be connected to any adapted voltage reference. For example, in the case (not shown) of a regulation by a measurement at the secondary, terminal  26  is connected to the transistor of an optocoupler (or the like) to provide a current control point. 
   The output of comparator  30  is connected to block  25  by a switch  28  controlled by a circuit  29  (TIME GEN), the function of which is to synchronize the turning-off of switch  28  with the beginning of each on cycle of switch  6 . This enables avoiding erroneous detection of current peaks at the turning-on. 
   A feature of the present invention is to use the result of a comparison of the current in switch  6  with a reference current to modify, if necessary, the switching frequency of the supply voltage. For this purpose, circuit  20  includes a second comparator  40 , a first input of which receives an information depending on the current in switch  6  and a second input of which receives a control point associated with a limiting current (block  41 , Ilim). Preferably, the control point corresponds to a fixed reference. The first input of comparator  40  is, for example, connected to an element  45  for measuring the source current in transistor  6  or exploits the result of element  31 . The output of comparator  40  is connected, by a switch  42 , to an integrator  43 , the output of which controls a current source  44  connected between terminal  24  and the ground. The function of the integrator is to accumulate the result of the comparison between two measurement periods set by the turn-on frequency of a switch  42 . Switch  42  is controlled by circuit  29 , which determines a delay with respect to the turning-on of switch  6  to turn on switch  42  and, further, determines the duration of conduction of switch  42 . 
     FIGS. 5A ,  5 B,  5 C,  5 D,  5 E, and  5 F illustrate, in the form of timing diagrams, the operation of a power converter according to the present invention, and more specifically of its control circuit.  FIG. 5A  shows an example of shape of control signal Vg of the gate of transistor  6 .  FIG. 5B  shows an example of shape of control signal Vb of switch  28 .  FIG. 5C  shows an example of shape of control signal Vs of switch  42 .  FIG. 5D  shows an example of current Id in switch  6 .  FIG. 5E  shows voltage Vi at the input of integrator  43 .  FIG. 5F  shows current  1  set by source  44 . The timing diagrams of  FIG. 5  show, in their left-hand portion, a normal operating cycle, and in their right-hand portions, a cycle in case of a short-circuit. 
   Assume a turning-on of switch  6  at a time t 1  ( FIG. 5B ). After a current peak (times t 1  to t 5 ) due to the switching, current Id progressively increases along the magnetization of the primary winding of the transformer until switch  6  is turned off (times t 3 ). The duration of an on pulse of switch  6  (times t 1  to t 3 ,  FIG. 5A ) generally is of a few microseconds (switching frequency greater than one kilohertz). 
   Control signal Vb of switch  28  is normally high, that is, switch  28  is in a normally on state. Circuit  29  turns it off for a short duration (a few hundreds of nanoseconds) after each turning-on of switch  6 . The duration of the low-state pulse of signal Vb, during which the control circuit does not take the comparison result of comparator  30  into account, is linked to the duration of the spurious current peak at the beginning of the magnetization cycle. The fact of not taking the spurious pulse into account has been illustrated in  FIG. 5D  by a plotting in dotted lines of the current peak at the beginning of the cycle. In the example of  FIG. 5B , signal Vb is low between times t 1  and t 5 . Interrupting the comparison enables avoiding activation of the current limiting means under the effect of spurious noise due to the switchings. 
   Switch  42  is in a normally off state, that is, signal Vs activates the turning-on when high. Thus, the result of the comparison provided by comparator  40  is only taken into account and integrated by circuit  43  during a short period in each on cycle of switch  6 . The aim is to compare the current of transistor  6  with the limiting current at the beginning of each on cycle. It must however be avoided to take spurious peaks into account. Accordingly, the turning-on of switch  42  occurs at time t 5  when the peak has disappeared. For example, this turning-on is simultaneous with the turning-on of switch  28 . The on-state duration of signal Vs is chosen to be short as compared to the on-state duration of switch  6 . For example, a duration on the order of a few hundreds of nanoseconds (times t 5  to t 4 ,  FIG. 5C ) enables taking into account the current at the beginning of the magnetization cycle. 
   The result provided by comparator  40 , more specifically at the input of integrator  43 , is illustrated in  FIG. 5E . In normal operation, the current increases from the origin, that is, a complete demagnetization is performed for each cycle. Accordingly, the result of the comparison does not indicate that the limiting current threshold has been exceeded. Current  1  in source  44  is then unmodified. 
   In case of a short-circuit (right-hand portion of  FIG. 5 ), the same time intervals are present. They are identified by the same references as in the left-hand portion of the timing diagrams, with an apostrophe (′). Since the secondary is short-circuited, the demagnetization is not complete. As a result, after a significant current peak (times t′ 1  to t′ 5 ), the current increase in the primary winding of the transformer does not start from zero but from a short-circuit value Icc. If this value is greater than current Ilim, the result provided by comparator  40  is positive and integrator  43  modifies the control of current source  44 . This results in an increase of current  1  from time t′ 5 . This increase is progressive under the effect of the pulse integration (times t′ 5  to t′ 4 ) performed by integrator  43 . From time t′ 4 , current  1  is constant until the next cycle. 
     FIG. 6  illustrates, with a different scale, three successive cycles where the secondary of the transformer remains in short-circuit. Current  1  increases by steps until reaching the maximum current of source  44 , set by construction. 
   The increase of current  1  sampled from terminal  24  decreases the frequency provided by block  27 . Accordingly, the switching frequency of switch  6  is decreased, which enables achieving again a complete demagnetization thereof. 
   The example shown in the right-hand portion of the timing diagrams of  FIG. 5  does not correspond to the cycle in which the short-circuit occurs. Indeed, a few cycles are necessary for the lack of demagnetization of the transformer to have it switch to a continuous mode. 
   The choice of the predetermined value Him of the current beyond which a lack of demagnetization is considered depends on the application and, in particular, on whether it is desired to allow or not for a continuous operating mode for the transformer. 
   By choosing a sufficiently low current Ilim, the switching to a continuous mode is forbidden. Indeed, current source  44  will be activated as soon as the current in switch  6  will not start from the origin. The chosen threshold however depends on the duration of the detection pulse. 
   According to a preferred embodiment of the present invention, the switching to a continuous mode of the transformer is allowed while limiting the current therethrough. This amounts to allowing for the continuous mode until a given threshold. In this case, the value of current Ilim is, preferably, between one quarter and three quarters (for example, one half) of the maximum current desired for the transformer. 
   An advantage of maintaining an operation in continuous mode is that the converter can then be used in bi-voltage applications. 
   An advantage of the present invention is that the provided solution is entirely integrable without requiring any additional terminal as compared to a conventional VIPER integrated circuit. 
   Another advantage of the present invention is that even in normal operation, that is, as long as maximum value Imax of the current set by comparator  30  has not been reached, the transformer is already protected by the choice of a limiting value Ilim smaller than this maximum value. 
   The influence of current source  44  may be adapted on the oscillation frequency by modifying the value of resistance R 3 . By choosing a relatively low resistance R 3 , an oscillation is guaranteed even when current source  44  has reached its maximum current. By choosing a relatively large resistance R 3 , the oscillation can stop, for example, when the value of current  1  has reached the maximum that the source can provide. 
   An advantage of the present invention thus is that a minimum switching frequency can be set. 
   Another advantage is that the starting of the power converter is then improved. Indeed, at the circuit starting, the capacitor (C 2 ,  FIG. 1 ) of the secondary of the transformer is entirely discharged. The converter then temporarily behaves as a short-circuit. The implementation of the present invention enables limiting the current for the first milliseconds of converter operation. 
   Of course, the present invention is likely to have various alterations, modifications, and improvement which will readily occur to those skilled in the art. In particular, the practical implementation of the comparison, integration, and current source functions are within the abilities of those skilled in the art based on the functional indications given hereabove. Further, the choice of the thresholds to be applied and of the component values depend on the application and on the authorization or not of the continuous operation mode, and will thus be adapted by those skilled in the art. 
   Such alterations, modifications, and improvements are intended to be part of this disclosure, and are intended to be within the spirit and the scope of the present invention. Accordingly, the foregoing description is by way of example only and is not intended to be limiting. The present invention is limited only as defined in the following claims and the equivalents thereto.