Abstract:
Techniques to adjust sampling times of an input signal. The techniques may utilize multi-level modification of the phase of a sampling clock. For example, the level of modification of the phase of the sampling clock may depend on the phase angle of the sampling clock in which transitions of the input signal occur.

Description:
FIELD 
   This invention generally relates to techniques for adjusting the horizontal sampling point of distorted data signals. 
   DESCRIPTION OF RELATED ART 
   Jitter is the general term used to describe distortion caused by short-term deviations of a signal from its ideal timing position. It is desirable to transmit signals having a minimal amount of jitter, but non-ideal components of a communication system inevitably distort the data signal. The distortion may be a deviation in the amplitude, time, or phase of the data signal and may cause errors in the recovery of data within a receiver. 
   Jitter may be divided into (1) random jitter (which can be caused by unavoidable noise sources within the system) and (2) deterministic jitter (which can be caused by non-ideal data signal transmission and processing). The random jitter component is considered to have a zero-mean gaussian (bell-shaped) distribution, whereas the deterministic jitter component may have both a discrete and a non-symmetric distribution. A conventional receiver may show degraded performance in cases where a non-gaussian jitter distribution is present. 
     FIGS. 1A–1C  depict different examples of (a) transmitted input signals shown in the form of eye-diagrams, (b) clock signals (labeled CLOCK) generated within the receiver and used to sample the received input signals, and (c) associated transition density diagrams of input signal. An eye diagram may represent the phases (i.e., time instances) of signal CLOCK at which transitions of the input signal occur. A transition of the input signal (i.e., a shift from ‘0’ to ‘1’ or from ‘1’ to ‘0’) may be defined as a crossing of the 50% level. 
   In some current systems, Alexander type phase comparators are used to regenerate the input signal. For a description of Alexander type comparators, see for example Electronic Letters by J. D. H. Alexander in an article entitled, Clock Recovery From Random Binary Signals, Volume 11, page 541–542, October 1975. In some systems, Alexander type comparators align the 180 degree phase of the CLOCK signal with the peak or the mean of the transition distribution and sample the input signal at each zero (0) degree phase of the CLOCK signal. Such samples of the input signal can be used to regenerate the input signal. 
     FIG. 1A  depicts a so-called “open eye” scenario with a non-return to zero (NRZ) input signal in which transitions of the input signal occur primarily within a narrow phase region (shown as region T). The transition distribution curve shows that transitions of the input signal occur primarily within a narrow phase region. When the 180 degree phase of the signal CLOCK is aligned with the peak of the transition distribution, the zero (0) degree phase of signal CLOCK may be used to sample the input signal. 
     FIG. 1B  depicts a distorted NRZ signal and a so-called “closed eye” scenario in which transitions of the input signal occur over a wide range of phases of signal CLOCK. Accordingly, the eye diagram of the input signal reveals a small region in which transitions do not occur. In this scenario, phase comparisons may indicate that transitions of the input signal frequently occur outside of region T. Region T may represent a region of contained transition distribution of the input signal. In this scenario, samples of the input signal taken at phase S may be taken when transitions of the input signal occur. If samples of the input signal are taken at phase S, the input signal may be improperly sampled. 
     FIG. 1C  depicts an asymmetric return to zero (RZ) signal scenario in which transitions of the input signal occur with a discrete distribution of phases of signal CLOCK. In this scenario, phase comparisons may indicate that the transition distribution of the input signal has several peaks with one peak at the center of region T. Furthermore, one of the edges of the input signal may be steeper and more well-defined than the opposite ‘blurred’ transition as depicted by the transition distribution function. In this scenario, the 180 degree phase of signal CLOCK may be aligned with one of the peaks of the transition distribution, and samples of the input signal taken at phase S may be made at a time when the “eye” is not completely open. Accordingly, samples of the input signal may be inaccurate. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIGS. 1A–1C  depict examples of a clock signal, an input signal, as well as associated transition density diagrams. 
       FIG. 2  depicts an example of a receiver system that may use some embodiments of the present invention. 
       FIG. 3  depicts in block diagram form an embodiment of the present invention in a sampling phase adjuster. 
       FIG. 4  depicts a sample transfer function of a phase comparator in accordance with an embodiment of the present invention. 
       FIG. 5  depicts an example implementation of a phase comparator in accordance with an embodiment of the present invention. 
       FIG. 6  depicts example transfer functions of leadA/lagA and leadB/lagB that phase comparators may implement, in accordance with an embodiment of the present invention. 
   

   Note that use of the same reference numbers in different figures indicates the same or like elements. 
   DETAILED DESCRIPTION 
   For example,  FIG. 2  depicts an example receiver system  20  that may use some embodiments of the present invention. Optical-to-electrical converter (“O/E”)  22  may convert optical signals received from an optical network into electrical signals. Although reference has been made to optical signals, the receiver  20  may, in addition or alternatively, receive electrical signals from an electrical signal network. Amplifier  24  may amplify the electrical signals. Re-timer system  25  may, in accordance with an embodiment of the present invention, adjust phases of a clock signal used to sample and reproduce an input signal. Re-timer system  25  may also regenerate electrical signals using the samples. On the regenerated signals, layer two processor  26  may perform media access control (MAC) management in compliance for example with Ethernet, described for example in versions of IEEE 802.3; optical transport network (OTN) de-framing and de-wrapping in compliance for example with ITU-T G.709; forward error correction (FEC) processing, in accordance with ITU-T G.975; and/or other layer  2  processing. Interface  28  may provide intercommunication between layer two processor  26  and other devices such as a switch fabric. Interface  28  may be compliant, for example, with a vendor specific multi-source agreement (MSA) protocol. The examples described with respect to  FIG. 1  by no means limit the systems in which some embodiments of the present invention may be used. For example, receiver  20  may be adapted to receive wireless or wire-line signals according to any standards. 
     FIG. 3  depicts in block diagram form an embodiment of the present invention in sampling phase adjuster  300 . Sampling phase adjuster  300  may adjust a sampling phase of an input signal (shown as signal INPUT) so that samples of signal INPUT may be accurately made. In accordance with an embodiment of the present invention, sampling phase adjuster  300  may adjust the sampling phase (e.g., zero (0) degree phase) of signal CLOCK to a phase angle where transitions of signal INPUT are less likely to occur. 
   For example, in the scenarios depicted in  FIGS. 1B and 1C , the sampling phase of signal CLOCK (phase S) corresponds to a phase where transitions of signal INPUT are likely to occur. In this example, in accordance with an embodiment of the present invention, sampling phase adjuster  300  may adjust the sampling phase of signal CLOCK from S to S′. Phase S′ may correspond to a phase where transitions of signal INPUT are less likely to occur than phase S and where the “eye-opening” may be largest (i.e., the phase in the eye diagram where the difference in magnitude between “1” and “0” values may be greatest). 
   One implementation of sampling phase adjuster  300  may include a clock generator  310 , phase comparator  320 , charge pump  330 , and loop filter  340 . Clock generator  310  may output a clock signal (shown as CLOCK). Clock generator  310  may adjust the phase of signal CLOCK based on control signal CNTRL. Clock generator  310  may be implemented as a voltage controlled oscillator (VCO) or voltage-controlled crystal oscillator (VCXO) although other oscillators may be used. 
   Phase comparator  320  may output samples of signal INPUT (such output samples are shown as signal OUTPUT) timed according to signal CLOCK. In one implementation, phase comparator  320  may sample signal INPUT at zero (0) degree phases of the signal CLOCK, although other phase angles may be used. In addition, phase comparator  320  may provide a signal (shown as lead/lag) to move the sampling phase of signal CLOCK. In one embodiment of the present invention, phase comparator  320  may move the sampling phase of signal CLOCK more when a transition of signal INPUT occurs within phase angles of signal CLOCK approximately between |±X| and 0 than when a transition of signal INPUT occurs within phase angles of signal CLOCK approximately between |±180| and |±X|. The value X may correspond to a phase angle of signal CLOCK where if transitions of signal INPUT approximately occur, the signal INPUT may have properties of a closed eye (such as described with respect to  FIGS. 1B and 1C ). A larger value of X may be chosen when a smaller transition region is desired. For example, in one implementation, the value X may be approximately 90 degrees although other values may be used. 
   For example,  FIG. 4  depicts a sample transfer function of a phase comparator  320  for a sampling phase of zero degrees and transition alignment at 180 degrees, in accordance with an embodiment of the present invention. In this example, if a transition of signal INPUT occurs among phase angles of signal CLOCK approximately between −180 and −X, phase comparator  320  may output signal lead/lag having a magnitude of −L to slow signal CLOCK by a phase amount proportional to L (which may correspond to a state where the signal INPUT lags the signal CLOCK). In this example, if a transition of signal INPUT occurs among phase angles of signal CLOCK approximately between −X and 0, phase comparator  320  may output signal lead/lag having a magnitude of −M, where M&gt;L, to slow signal CLOCK by a phase amount proportional to M (which may correspond to a state where the signal INPUT lags the signal CLOCK). 
   In this example, if a transition of signal INPUT occurs among phase angles of signal CLOCK approximately between X and 180 degrees, phase comparator  320  may output signal lead/lag having a magnitude of L to speed signal CLOCK by a phase amount proportional to L (which may correspond to a state where the signal INPUT leads the signal CLOCK). In this example, if a transition of signal INPUT occurs among phase angles of signal CLOCK approximately between 0 and X degrees, phase comparator  320  may output signal lead/lag having a magnitude of M, where M&gt;L, to speed signal CLOCK by a phase amount proportional to M (which may correspond to a state where the signal INPUT leads the signal CLOCK). 
     FIG. 5  depicts an example implementation of phase comparator  320  in accordance with an embodiment of the present invention. Phase comparator  320  may include phase comparators  510 A and  510 B and adder  520 . For example,  FIG. 6  depicts example transfer functions of signals leadA/lagA and leadB/lagB that respective phase comparators  510 A and  510 B may implement. 
   For 0, 180, and −180 degree phase angles of signal CLOCK, phase comparator  510 A may compare the transitions of signal INPUT with transitions of a signal CLOCK and indicate whether the transitions of the signal INPUT lead or lag those of signal CLOCK (such output shown as leadA/lagA). Herein, signal CLOCK−X may represent a version of signal CLOCK phase shifted by −X degrees. Herein, signal CLOCK+X may represent a version of signal CLOCK phase shifted by +X degrees. For X and −X degree phase angles of signal CLOCK, phase comparator  510 B may compare the transitions of signal INPUT with transitions of respective signals CLOCK+X and CLOCK−X and indicate whether the transitions of the signal INPUT lead or lag transitions of respective signals CLOCK+X and CLOCK−X (such output shown as leadB/lagB). The signal lead/lag (having a transfer function depicted in  FIG. 4 ) may represent a sum of signals leadA/lagA and leadB/lagB. 
   Each of phase comparators  510 A and  510 B may be implemented as Alexander (“bang-bang”) type circuits. One possible implementation of an Alexander phase detector is described in Electronic Letters by J. D. H. Alexander in an article entitled, Clock Recovery From Random Binary Signals, Volume 11, page 541–542, October 1975. 
   Phase comparators  510 A and  510 B may output signals leadA/lagA and leadB/lagB to adder  520 . Adder  520  may sum signals leadA/lagA and leadB/lagB and provide the sum as signal lead/lag to charge pump  330  ( FIG. 3 ). 
   Referring to  FIG. 3 , charge pump  330  may receive signal lead/lag from phase comparator  320 . Charge pump  330  may add or remove charge from clock generator  310  in an amount in proportion to the sign (positive or negative) and magnitude of signal lead/lag. Charge pump  330  may output signal CNTRL that instructs the clock generator  310  to either increase or decrease the speed of signal CLOCK. For example, if charge pump  330  receives a lead indicator, signal CNTRL may correspond to charge addition to the clock generator  310  to increase the speed of the signal CLOCK. Conversely, if charge pump  330  receives a lag indicator, signal CNTRL may correspond to removal of charge from the clock generator  310  to decrease the speed of the signal CLOCK. 
   Loop filter  340  may transfer signal CNTRL to clock generator  310  when the frequency of signal CNTRL is within the pass band of the loop filter  340 . Clock generator  310  may receive the transferred portion of the sum of signal CNTRL. Although a charge pump and loop filter combination is provided as an example implementation, other devices may be used to selectively transfer signal CNTRL to the clock generator  310 . 
   The drawings and the forgoing description gave examples of the present invention. The scope of the present invention, however, is by no means limited by these specific examples. Numerous variations, whether explicitly given in the specification or not, such as differences in structure, dimension, and use of material, are possible. The scope of the invention is at least as broad as given by the following claims.