Abstract:
The two-stage power amplifier includes: a first stage transconductor  60 ; and a second stage having at least two parallel output branches  57-59  supplying current to an output node  89 , each output branch has an input coupled to an output of the first stage transconductor  60.

Description:
This application claims priority under 35 U.S.C. §119 (e)(1) of provisional application No. 60/151,141, filed Aug. 27, 1999. 
    
    
     FIELD OF THE INVENTION 
     This invention generally relates to electronic systems and in particular it relates to analog amplifier circuits. 
     BACKGROUND OF THE INVENTION 
     A typical prior art 2-stage amplifier block diagram is shown in FIG.  1 . The circuit of FIG. 1 includes transconductors  20  and  22 , resistors  24 - 26 , capacitors  28 - 30 , output resistance  32 , output capacitance  34 , input voltage V in , and output voltage V out . The prior art circuit of FIG. 2 shows the last section of a prior art folded cascode input stage coupled to an output stage as used in transconductors  20  and  22 . The circuit of FIG. 2 includes output transistors  40  and  42 , capacitors  44  and  46 , quiescent current bias network  48 , output voltage V out , and transistors  50 - 53 . To get maximum current drive, transistors  40  and  42  need to be very large devices. In order to not waste quiescent current, the gate-to-source voltage minus the threshold voltage (V GS −V T ) of transistors  40  and  42  are set as small as possible. The maximum output current determines the size of transistors  40  and  42 . The quiescent current is set by the second stage transconductance and the total harmonic distortion (THD) level of performance at small signal levels. At very low V GS −V T  the quiescent current is less controlled as transistors  40  and  42  head for subthreshold region of operation. Another problem with very low V GS −V T  is that the input stage cascodes may not have the head room required for maximum gain out of the stage. This is certainly true for modern CMOS processes where the V T  of the devices are going down, and operating voltages for systems is going down as well. Another problem is the very large parasitic capacitance of the large output transistors  40  and  42 . Typical load resistances for designs using CMOS power amplifiers are 32 Ohm all the way down to 8 ohm. With signal swings in the 4 volt range, this translates to current in the 70 mA to 250 mA range without sacrificing performance. These large power levels even with 90 Angstrom gate oxide on analog processes can result in PMOS devices sizes approaching 20,000 um (W/L). In these cases the parasitic capacitance of the output transistors  40  and  42  would be enormous, and would cause major problems with the stability of the amplifier. Using these prior art techniques, the quiescent current needed for proper operation of an amplifier of this type is very high. An amplifier of this type achieves good overall performance only at a maximum current to quiescent current ratio of 60-70. For proper operation the quiescent current level has to be such that the maximum load current is about 60-70 times the quiescent current. This does not solve the stability issue of the parasitic capacitance being very large. 
     SUMMARY OF THE INVENTION 
     A two-stage power amplifier includes: a first stage transconductor; and second stage having at least two parallel output branches supplying current to an output node, each output branch has an input coupled to an output of the first stage amplifier. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     In the drawings: 
     FIG. 1 is a typical prior art two-stage amplifier block diagram; 
     FIG. 2 is a schematic circuit diagram of the last section of a prior art folded cascode input stage coupled to an output stage; 
     FIG. 3 is diagram of a preferred embodiment two stage power amplifier with multiple output branches. 
     FIG. 4 is a partial detailed circuit diagram of the preferred embodiment of FIG. 3 with a single output branch. 
     FIG. 5 is a partial detailed circuit diagram of multiple source follower output stages. 
    
    
     DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS 
     The preferred embodiment circuit of FIG. 3 is a two-stage power amplifier with multiple output branches. The circuit of FIG. 3 includes first stage transconductor  60 , unity gain amplifiers  62 - 64 , second stage transconductors  66 - 68 , resistors  70  and  72 , capacitors  74  and  76 , Miller capacitor  78 , capacitors  80 - 82 , load resistance  84 , load capacitance  86 , common node  88 , output node  89 , input voltage V IN  and output voltage V out . The circuit of FIG. 3 is a two-stage design compensated by Miller capacitor  78 . The transconductance of the second stage of the circuit of FIG. 3 is the sum of the transconductance of all the branches. The unity gain amplifiers  62 - 64  level shift the bias point for each branch. The circuit of FIG. 3 is shown with three second stage branches  57 - 59 , but any number of branches could be used. 
     FIG. 4 is a partial detailed circuit diagram of the preferred embodiment of FIG.  3 . The circuit of FIG. 4 includes NMOS transistors  90 - 97 , PMOS transistors  99 - 101 , current sources  102  and  104 , and output voltage V out . Circuit  106  is the output section of the folded cascode input stage  60 , shown in FIG.  3 . The biasing technique (class AB) that includes transistors  90 ,  92 ,  95 , and  97  is modified from the prior art designs by the addition of transistors  91  and  96 . This moves node  108  lower in voltage so that the cascode can have more headroom which was one of the problems of the prior art device of FIG.  2 . The voltage at node  110  still remains equal to the voltage at node  112 , so that current I l  multiplied by the width-to-length (W/L) ratio of transistor  97  to transistor  90  flows through transistor  97 . The circuit of FIG. 4 also isolates the input stage from the very large parasitic capacitance of transistor  97  with the source-follower formed by transistor  96 . The large parasitic capacitance is then easier to drive. Transistor  96  performs as the DC level shifter for the biasing and as an AC type source-follower. 
     To achieve the desired current ratio of the output current to the quiescent current, the circuit of FIG. 4 is extended as shown by the circuit of FIG.  5 . The circuit of FIG. 5 includes source follower transistors  120 - 122 , output transistors  124 - 126 , current sources  128 - 130 , input node  132 , output node  134 , and output voltage V out . By taking multiple source-follower output device branches biased at different points, multiple large drivers are put in parallel, but not all of them are conducting large currents at quiescent. They can be throttled back to have a small current when desired. 
     In the circuit of FIG. 5, transistor  124  is the same as transistor  97  in FIG.  4 . Transistors  121  and  122  are sized smaller than transistor  120 , but conduct the same current. Because the (V GS −V T ) of transistors  121  and  122  is greater than for transistor  120 , the bias point for the output transistors  125  and  126  will be higher. Transistors  125  and  126  are as large as or larger than transistor  124 , which allows transistors  125  and  126  to be pushed into subthreshold operation. In this state, they conduct very small currents which improves the quiescent current value. This gives the amplifier much more current drive for maximum signal peaks. A much higher total output current capacity is provided. Because all of the branches are biased from a dependent node  132 , the biasing scheme still holds. If multiple branches are added with independent sections, it looks like two or more different gain paths to the output node and the feedback around the amp will not allow this in closed-loop operation. 
     One advantage of the preferred embodiment is that it solves the problem of huge parasitics caused by large power devices. The source follower stages distribute the massive output device into many branches in parallel. This reduces the parasitic capacitance to a more reasonable size. Another advantage is that all the additional output branches that are in subthreshold burn little current, but their transconductance is not small. These branches do contribute much more total transconductance to the second stage (the transconductances sum to the output node), therefore the DC gain into such a small load resistance is much improved. Also, the compensation capacitor C c  can be smaller. 
     Although the present invention has been described in detail, it should be understood that various changes, substitutions and alterations can be made without departing from the spirit and scope of the invention as defined by the appended claims. It is therefore intended that the appended claims encompass any such modifications or embodiments.