Abstract:
A method and apparatus for reducing the size of microwave (or millimeter wave) dielectric resonator filters and for tuning the filter by inserting tuning screw within the dielectric itself. The filter includes a metallic housing that encloses a plurality of cavities, and each cavity contains a dielectric resonator whose top and bottom surfaces are flush with the top and bottom walls of the metallic structure. Due to the continuity and uniformity of the electric field generated in the y-axis of the dielectric, the filter&#39;s performance response becomes independent of height. This novel design allows for substantial reduction in cavity size without appreciably dropping the Q factor. Such continuity and uniformity of the electric field also allows for openings to be made parallel to the y-axis and inside the dielectric resonator, wherein tuning screws are inserted to selectively adjust the frequency. Other aspects of the invention include alternative methods for electromagnetic coupling in, within, and out of the filter; methods for reducing the machining accuracy by creating a small air gap at one end of the resonator; and methods for reducing the propagation of high modes by alternating the shapes or orientation of the resonators within the filter.

Description:
FIELD OF THE INVENTION  
         [0001]    The present invention relates to microwave filters in wireless telecommunications systems. In particular, the present invention relates to dielectric resonator filters operating in microwave and millimeter wave rectangular waveguides or cavities of transceivers.  
         BACKGROUND OF THE INVENTION  
         [0002]    Over the years a wide variety of microwave and millimeter wave filters have been developed, each satisfying specific application requirements but none offering the optimum combination of low insertion loss, higher order mode rejection, high unloaded Q factor, high temperature stability, reduced filter size, tunability, and ease of manufacturing.  
           [0003]    The first-generation filters consisted of empty cascaded conductive cavities connected together and separated by metallic walls with iris-controlled couplings. These filters are bulky and not particularly suitable for use at low frequencies such as those below the X-band. One solution to this problem was the construction of a coaxial structure supporting a TEM mode with a capacitive gap called a comb-line, as described in G. L. Matthaei, “Comb-line Bandpass Filters of Narrow or Moderate Bandwidth”, Microwave Journal, Vol. 6, August 1963. While this technology offers a greater reduction in size compared to the size of empty rectangular or cylindrical cavities, its moderate Q factor does not meet the stringent Q factor specifications required in certain modern telecommunication systems.  
           [0004]    To obtain a high Q factor, the filter configurations most commonly used in today&#39;s telecommunication systems consist of a dielectric puck mounted inside a conductive housing without touching the metal conductor, as described in the following references: (a) J. F. Liang and W. D. Blaire, “High Q TE 01  Mode DR Filters for PCS Wireless Base Stations”, IEEE Transactions, Microwave Theory Tech., Vol. 1, MTT-46, Dec. 1998; (b) X-P Liang and K. A. Zaki, “Modeling of Cylindrical Dielectric Resonators in Rectangular Waveguides and Cavities”, IEEE Trans. Microwave Theory Tech., Vol. MTT-41, Dec. 1993: and (c) U.S. Pat. No. 5,777,534 to Harrison et al., entitled “Inductor Ring for Providing Tuning and Coupling in Microwave Dielectric Resonator Filters”. In these structures the electromagnetic field is concentrated inside the puck and vanishes gradually outside. While the relatively wide cavity used in these structures reduces the ohmic loss on the metallic wall and increases the Q factor, it also increases the size and weight of the filter. Moreover, an undesirable electromagnetic mode (called the HE mnδ  mode) is excited in such structures. This mode produces spurious responses close to the filter bandwidth, which affects the filter rejection performance.  
           [0005]    With the advent of cellular mobile phone systems, new filter technologies using dielectric materials have been developed which yield moderate Q factors and reduced size, such as that described in Kikuo Wakino et al, “Miniaturization Technologies of Dielectric Resonator Filters for Mobile Communications”, IEEE Trans. Microwave Theory Tech., Vol. MTT-42, July 1994. However, the topology of the majority of these technologies involve complex geometry that requires high machining accuracy and increased assembly time.  
           [0006]    Other recent technologies have been developed to reduce spurious response. A simple configuration of such schemes has been proposed by A. Abdelmonem, J-F. Liang and K. A. Zaki, “Full-wave Design of Spurious-free DR TE Mode Bandpass Filters”, IEEE Trans. Microwave Theory Tech., Vol. MTT-43, April 1995. While the spurious response in this structure is substantially free, the resonators are not tunable. They also require high machining tolerance and high precision in the selection of the value of the dielectric constant.  
           [0007]    An example of a prior art device tuning arrangement for a dielectric resonator filter  40  is illustrated in FIG. 1. The filter  40  includes a metallic disk  42  attached to the upper surface of a housing structure  44  by a screw  46 . A dielectric resonator  48  is mounted on a support  50  centrally positioned within a cavity  52  of filter  40 . The distance between the top surface of the resonator  48  and the bottom surface of the disk  42  can be varied up and down by rotating the screw  46 . The disk  42  interacts with the magnetic field of the resonator  48  causing perturbation of the resonance frequency of the cavity  52 . A disadvantage of this Topology is the excitation of undesirable spurious hybrid modes at frequencies that are close to the filter&#39;s passband.  
           [0008]    It is therefore desirable to provide a substantially smaller-size filter for both microwave and millimeter wave frequency bands that uses internally-tunable dielectric resonators. It is further desirable to provide dielectric resonators that have a high Q factor, are easily manufactured and mounted, and provide substantial improvement in out-of-band hybrid mode rejection performances.  
         SUMMARY OF THE INVENTION  
         [0009]    It is an object of the present invention to obviate or mitigate at least one disadvantage of prior art bandpass filters. In particular, it is an object of the present invention to provide a dielectric resonator filter, particularly for microwave and millimeter wave applications, that is tunable.  
           [0010]    In accordance with a first aspect of the present invention , there is provided a tunable dielectric resonator filter. The tunable dielectric resonator filter consists of an electrically conductive housing defining a cavity, and a dielectric resonator disposed in the cavity. A tuning aperture is formed in the resonator. The aperture is substantially parallel to a direction of an electric field excited within the resonator. A tuning device, such as a rod or screw, received within the tuning aperture. The depth of penetration of the tuning device within the resonator determines a frequency response of the resonator.  
           [0011]    Typically, a coupling probe is provided to couple a signal to and from the cavity. The coupling probe excites the cavity in a TE mode, and can be within the cavity or disposed in a coupling aperture provided in the resonator. The filter of the present invention in effectively excited in a LSE mode. The resonator can be provided with an electrically conductive coating, on any of its top, bottom or side surfaces.  
           [0012]    By coupling together a series of dielectric resonator filters according to the present invention, a tunable bandpass filter can be formed. Typically, the coupling is achieved by irises. Alternatively, an oscillator can be formed by coupling together a dielectric resonator filter according to the present invention with an oscillating element. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0013]    Preferred embodiments of the present invention will now be described, by way of example only, with reference to the attached figures wherein:  
         [0014]    [0014]FIG. 1 is a side view of a prior art filter;  
         [0015]    [0015]FIG. 2 is a top view of a six-pole, dielectric resonator filter in accordance with the present invention;  
         [0016]    [0016]FIG. 3 is a cross-sectional view of the dielectric resonator filter shown in FIG. 2;  
         [0017]    [0017]FIG. 4 is a top view of a filter cavity showing the unloaded and loaded sections of a rectangular resonator;  
         [0018]    [0018]FIG. 5 is a top view of a filter cavity showing the unloaded and loaded sections of a cylindrical resonator;  
         [0019]    [0019]FIG. 6 is a cross-sectional view of FIG. 4 or FIG. 5 showing the uniformity of the dielectric resonator geometry in the direction of the electric field;  
         [0020]    [0020]FIG. 7 is a cross-sectional view of the input/output coupling section of a filter having a shorted coupling rod positioned outside the dielectric resonator in accordance with the present invention;  
         [0021]    [0021]FIG. 8 is a cross-sectional view of the input/output coupling section of a filter having an open-ended coupling rod positioned outside the dielectric resonator in accordance with the present invention;  
         [0022]    [0022]FIG. 9 is a cross-sectional view of the input/output coupling section of a filter having an open-ended coupling rod positioned within the dielectric resonator in accordance with the present invention;  
         [0023]    [0023]FIG. 10 is a cross-sectional view of a filter having two open-ended cross-coupling rods between two non-adjacent dielectric resonators in accordance with the present invention;  
         [0024]    [0024]FIG. 11 is a perspective view of a dielectric resonator inserted in a rectangular metallic housing in accordance with the present invention;  
         [0025]    [0025]FIG. 12 is a perspective view of a dielectric resonator inserted in a rectangular metallic housing showing a small air gap between the top of the resonator and the top of the housing;  
         [0026]    [0026]FIG. 13 is a cross-sectional view of a dielectric resonator inserted in a rectangular metallic housing showing the insertion of an expandable conductor slab in the air gap of FIG. 12;  
         [0027]    [0027]FIG. 14 is a perspective view of a rectangular dielectric resonator that has been metal-plated on its top and bottom surfaces;  
         [0028]    [0028]FIG. 15 is a perspective view of a rectangular dielectric resonator that has been metal-plated only on its bottom surface in accordance with another aspect of the present invention.  
         [0029]    [0029]FIG. 16 is a perspective view of a cylindrical dielectric resonator that has been metal-plated on its top and bottom surfaces;  
         [0030]    [0030]FIG. 17 is a perspective view of a cylindrical dielectric resonator that has been metal-plated only on its bottom surface;  
         [0031]    [0031]FIG. 18 is a top view of a filter showing the longer-spaced coupling between two adjacent rectangular resonators without an iris coupler;  
         [0032]    [0032]FIG. 19 is a top view of a filter showing the longer-spaced coupling between two adjacent cylindrical resonators without an iris coupler;  
         [0033]    [0033]FIG. 20 is a top view of a filter showing the shorter-spaced coupling between two adjacent rectangular resonators with an iris coupler;  
         [0034]    [0034]FIG. 21 is a top view of a filter showing the shorter-spaced coupling between two adjacent cylindrical resonators with an iris coupler;  
         [0035]    [0035]FIG. 22 is a perspective view of a rectangular resonator with partial metallic plating on one of its lateral sides;  
         [0036]    [0036]FIG. 23 is a perspective view of a cylindrical resonator with partial metallic plating on its cylindrical surface;  
         [0037]    [0037]FIG. 24 is a top view of a filter showing rectangular and cylindrical resonators adjacent to one another;  
         [0038]    [0038]FIG. 25 is a top view of a filter showing two similar rectangular resonators positioned 90° from one another;  
         [0039]    [0039]FIG. 26 is a graph showing the measured insertion loss and return loss responses of a reduced-size filter constructed in accordance with the present invention; 
     
    
     DETAILED DESCRIPTION OF THE INVENTION  
       [0040]    Generally, the present invention provides a tunable dielectric resonator filter operating in a LSE 10δ  mode. The filter of the present invention is substantially reduced in size and weight when compared to prior art TE 01δ  filters. Further, it is much easier to tune than prior art dielectric resonator filters, while still satisfying the desired requirements of low insertion loss, good out-of-band rejection performance, relatively large unloaded Qs, high-temperature stability, and ease of manufacturing and mounting.  
         [0041]    Referring now to FIG. 2 and FIG. 3, there is shown a top view and a cross-sectional view of a six-pole, dielectric resonator filter  60  according to one aspect of the present invention, including six resonant cavities  62 ,  64 ,  66 ,  68 ,  70  and  72  housed within the metallic walls of a rectangular waveguide structure  74 . External coupling of the filter is performed by the coupling devices  76 ,  78  and  80 , 82 , whereas internal coupling between cavities is performed by the irises  84 ,  86 ,  88 ,  90 , and  92  and by the cross coupler  94 . Rectangular-shaped dielectric resonators  96 ,  98 ,  100 ,  102 ,  104  and  106 , having a high dielectric constant and high intrinsic Q, are positioned centrally within their respective cavities and flush with the top and bottom walls of the metallic structure  74 , as shown in FIG. 3. Substantially central to each dielectric resonator and in the same direction as the electric field (y-axis) is an opening that penetrates the entire resonator, allowing for the insertion of metallic or dielectric tuning screws (or rods)  108 ,  110  and  112 .  
         [0042]    Noted that no relative dimensional information should be inferred from these figures, that a smaller or greater number of cavities may be used according to the frequency selectivity requirements of the filter and according to the teachings of the present disclosure, and that alternative forms or shapes of the dielectric resonator, such as puck-shaped disks, may be used. Considering now the structural configuration of the preferred embodiment of FIG. 2, the present invention will be described by way of the electromagnetic signal that propagates through the cavities and by showing how certain characteristics of the derived equations allow for a wide range of trade-off possibilities between the Q factor and the structural dimension.  
         [0043]    Due to the geometry of the metallic waveguide structure  74  and the orientation of the coupling probe  82  of FIG. 3, the signal propagating in the unloaded section of the cavity (as shown at  118  of FIGS. 4, 5 and  6 ), operates in the standard TE 01  mode. With the common factor e jwt  removed, the components of the electromagnetic field of the signal are given by the super-positioning of incoming and reflected TE no  modes as follows:  
         E   y   I     =         ∑   n            F   n   I          φ   n                 -     γ   n          z           +       ∑   n            B   n   I          φ   n                 γ   n        z                       H   x   I     =       j     ωμ   0            [         ∑   n            F   n   I          γ   n          φ   n                 -     γ   n          z           -       ∑   n            B   n   I          γ   n          φ   n                 γ   n        z             ]                 H   z   I     =       j     ωμ   0            [         ∑   n            F   n   I          φ   n   ′                 -     γ   n          z           +       ∑   n            B   n   I          φ   n   ′                 γ   n        z             ]             where             γ   n     =           (       n                 π     a     )     2     -       ω   2          μ   0          ɛ   0             ,     
            φ   n     =       cos                   (         n                 π     a        x     )                   and                   φ   n   ′       =       ∂     φ   n         ∂   x                                 
 
         [0044]    However, as the signal propagates through the loaded section of the cavity, the components of the electromagnetic field are altered due to the super-positioning of the incoming and reflected LSE mo  modes. In the section loaded with a rectangular dielectric resonator (as shown at section  120  of FIG. 4), the components of the electromagnetic field are given by the following equations:  
         E   y   II     =         ∑   m            F   m   II          ψ   m                 -     Γ   m          z           +       ∑   m            B   m   II          ψ   m                 Γ   m        z                       H   x   II     =       j     ωμ   0            [         ∑   m            F   m   II          Γ   m          φ   m                 -     Γ   m          z           -       ∑   m            B   m   II          γ   m          ψ   m                 Γ   m        z             ]                 H   z   II     =       j     ωμ   0            [         ∑   m            F   m   II          ψ   m   ′                 -     Γ   m          z           -       ∑   m            B   m   II          ψ   m   ′                 Γ   m        z             ]             where           ψ   m   ′     =       ∂     ψ   m         ∂   x                         ψ   m     =       sin              [       χ     1      m            (       a   -   d     2     )       ]        cos                   (       χ     2      m          x     )                              for                 x     &lt;     d   2                   ψ   m     =       cos        [       χ     2      m            (     d   2     )       ]            sin              [       χ     1      m            (       a   2     -   x     )       ]                 for                 x     &gt;     d   2                                 
 
         [0045]    Similarly, in a section loaded with a cylindrical dielectric resonator (as shown at  121  of FIG. 5) the components of the electromagnetic field are given by the following equations:  
         E   y   II     =       ∑   m            F   m   II            Z   m          (   kr   )          cos                   (     m                 θ     )                   H   x   II     =         -   j       ωμ   0              ∑   m            n   r          F   m   II            Z   m          (   kr   )          sin                   (     m                 θ     )                     H   z   II     =         -   j       ωμ   0              ∑   m            F   m   II            kZ   m   ′          (   kr   )          cos                   (     m                 θ     )                                 
 
         [0046]    where  
           Z   m ( kr )= f   m   J   m ( kr )+ Y   m ( kr )  
         [0047]    is a linear combination of Bessel and Neumann functions of the order n.  
         [0048]    In the second and third sets of the above equations (for the loaded sections), the values of the constants X 1m , X 2m , γ m  and F m  are generally obtained by satisfying the continuity conditions of the field on the air/dielectric interfaces and the boundary conditions of the lateral conductor walls. While these parameters vary according to the cavity width, the permitivity of the loaded section, and the dielectric resonator width, they do not depend on the resonator height. It follows therefore that, due to the uniformity of the electric field in the y axis (as shown in FIG. 6), the performance response of the filter regarding the central frequency, bandwidth, and return loss is not affected by changing the height of the filter. Thus, the structural configuration of the present invention (FIG. 2) allows for a wide range of trade-off selections between the Q factor and the filter dimension, and it can be shown that, while remaining well within the imposed selectivity limits, a nominal drop in the Q factor can result in an appreciable reduction in resonator size. This characteristic feature of height independence along the y-axis of tunable dielectric resonators is unique to the present invention.  
         [0049]    Considering again the structural configuration of the presently preferred embodiment of the present invention (FIG. 2), it can be seen that the resulting uniformity of the electrical field along the y-axis allows for holes  122 ,  124  and  126  to be bored parallel to the y-axis and substantially central to, and within, the dielectric resonators. Said holes allow for the insertion of conductive or dielectric screws (or rods)  108 ,  110  and  112 . Upward or downward adjustment of these tuning devices causes perturbation of the electric field distribution E y   II  of the mode propagating within the respective resonators which, in turn, allows for an appreciable shift in frequency and good tuning of the filter. This internal method for tuning the dielectric resonator is unique to this invention.  
         [0050]    Additional tuning of the filter is also made possible under the preferred embodiment as shown in FIG. 3. The tuning devices  128  and  130  are positioned centrally between adjacent dielectric resonators. Upward or downward adjustment of these tuning devices causes perturbation of the electromagnetic field distribution in the TE n0  mode propagating between the resonators which, in turn, allows for tuning of the filter.  
         [0051]    In the preferred embodiment of the present invention the input and output coupling, shown in the unloaded sections  62  and  72  of FIG. 2 and FIG. 3, are performed by a shorted rod  78  or  82  as shown in FIG. 7, or by an open rod  132  as shown in FIG. 8. Since this coupling occurs below the cut-off region of the waveguide section, it has less coupling efficiency. This coupling method is better suited for narrow band filter applications.  
         [0052]    However, in accordance with another aspect of the present invention, a stronger coupling is made possible for wider band filter applications by inserting the coupling rod  134  through a hole  136  within the dielectric resonator, as shown in FIG. 9. This coupling method is much more efficient than those shown in FIG. 7 and FIG. 8 because the coupling rod  134  is positioned substantially within the concentrated portion of the electrical field.  
         [0053]    In yet another embodiment of the present invention, a dual probe  94  is inserted between two non-adjacent dielectric resonators, as shown in FIG. 10. Due to the available space between the dielectric resonator and the lateral wall of the filter, the insertion of a probe within said open space allows for negative cross-coupling between the two non-adjacent resonators. To avoid shorting, the probe  94  is isolated by the dielectric material  138 . Additionally, the resonator cross-coupling can be made tunable by connecting the probe  94  to a tuning screw  140 , as shown in FIG. 10. Upward or downward adjustment of the tuning screw causes a change in probe position between the two non-adjacent resonators, which, in turn, alters the cross-coupling.  
         [0054]    Alternatively, positive cross-coupling between the two non-adjacent dielectric resonators can be achieved by simply opening a small iris in the lateral wall facing the two non-adjacent resonators.  
         [0055]    In the presently preferred embodiment of the present invention, the top and bottom of the resonators are in perfect contact with the top and bottom walls of the waveguide structure  74 , as shown in FIG. 11. The key advantages of this aspect of the invention are that (a) it avoids propagation of spurious hybrid modes within the filter, (b) it permits reduction in filter size (height independence), and (c) it provides for good thermal conductivity. To achieve a good contact between the resonator and the waveguide walls, the top and bottom of the resonator are plated with a conductive material such as silver or copper or other metallic material, as shown by the metal strips  146  and  148  of FIG. 14 and FIG. 16.  
         [0056]    The disadvantage of the tight-fitting configuration of FIG. 11 is that it requires high machining accuracy. To reduce this constraint in topology, an alternative embodiment of the present invention is proposed by introducing a small air gap  142  between the top of the dielectric resonator and the top wall of the waveguide structure  74 , as shown in FIG. 12. For a small gap, the equations given above remain basically unaltered if the permitivity is changed by the effective corrective value, and the propagated mode in the loaded section merely changes from a pure LSE mode to a quasi LSE mode. Thus, for the same frequency application, the drawback resulting from this alternative embodiment is a slight increase in the width of the dielectric resonator and the introduction of a small amount of hybrid mode propagation. However, in accordance with a further aspect of the present invention, this drawback can be rectified by filling the air gap  142  with an expandable conductive slab  144 , as shown in FIG. 13.  
         [0057]    In the presently preferred embodiment of the present invention, the coupling distance between adjacent dielectric resonators can be reduced by the classic prior art method of inserting irises  150  or  152  between rectangular dielectric resonators  151  or cylindrical dielectric resonators  153 , as shown in FIG. 20 and FIG. 21. FIGS. 18 and 19 show respective dielectric resonators  151  and  153  without coupling irises. In single-mode filter designs, such a coupling method is required in order to reduce the otherwise wide spacing between adjacent resonators. In yet another aspect of the present invention, it is proposed to reduce the coupling distance between resonators even further by partially plating one lateral face  154  or  156  of the dielectric block with silver, copper, or other metallic material, as shown in FIG. 22 and FIG. 23.  
         [0058]    In accordance with yet another aspect of the present invention, it is proposed to use different resonator shapes  151  and  153  or to rotate adjacent resonators  900  from one another, as shown in FIG. 24 and FIG. 25. Depending on the permitivity, dimension, and/or shape of the dielectric resonator, the second mode LSE 201  can vary between 1.2 and 2.5 times the “central frequency” of the filter. Therefore, by changing the configuration of the resonators as shown in FIG. 24 or FIG. 25, the propagation of this mode can be substantially reduced.  
         [0059]    [0059]FIG. 26 shows the measured frequency response of a reduced-size filter constructed in accordance with the preferred embodiment of the present invention (FIG. 2). The two s-parameter curves illustrate the excellent performance of the filter in comparison with the larger-sized comb-line or cylindrical-puck dielectric filters of the prior art.  
         [0060]    As will be understood by those of skill in the art, the present invention provides the ability to tune a dielectric resonator filter operating in a LSE 10δ  mode by the simple expedient of tuning screws or rods. The present invention can provide either positive or negative tunable cross-coupling between at least two non-adjacent dielectric resonators in a rectangular waveguide filter. Ideally, the dielectric resonators of the present invention are flush with the upper and lower walls of the metallic waveguide housing. However, by removing the metal from one of the resonator&#39;s surface and introducing a small air gap between the top of the dielectric resonator and the top wall of the waveguide structure, the manufacturing and mounting process can be simplified without compromising performance. Further, the coupling distance between adjacent dielectric resonators can be significantly reduced by partially plating one adjacent face of the dielectric block with conductive metallic material. Equally, enhanced performance can be achieved by using different resonator shapes or rotating adjacent resonators 90° from one another in order to reduce the propagation of spurious hybrid modes.  
         [0061]    The above-described embodiments of the invention are intended to be examples of the present invention. Alterations, modifications and variations may be effected in the particular embodiments by those skilled in the art, without departing from the scope of the invention which is defined solely by the claims appended hereto.