Abstract:
A multiple-phase power converter comprises a non-isolated, double-ended transformer having a plurality of windings, a high-side switch portion, a low-side switch portion, an output portion, and a controller. The high-side switch portion includes a first power switch connecting an input voltage source (V IN ) to a virtual phase node through a first winding of the plurality of windings and a second power switch connecting the input voltage source to the virtual phase node through a second winding of the plurality of windings. The first and second windings are arranged with opposite polarity. The low-side switch portion includes a third power switch connecting the virtual phase node to ground through a third winding of the plurality of windings and a fourth power switch connecting the virtual phase node to ground through a fourth winding of the plurality of windings. The third and fourth windings are arranged with opposite polarity. The output portion includes an output inductor connecting the virtual phase node to an output terminal providing an output voltage (V OUT ). The controller is adapted to control operations of the first, second, third and fourth power switches such that the first and second power switches are enabled in respective alternating phases. The third switch is disabled when the second switch is enabled so that current flows concurrently through both the second and fourth windings to the output inductor during a first phase, and the fourth switch is disabled when the first switch is enabled so that current flows concurrently through the first and third windings to the output inductor.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to voltage regulator circuits. More particularly, the invention relates to a multi-phase power converter that can operate at high switching frequencies with effective ripple cancellation and reduced switching loss. 
     2. Description of Related Art 
     Switched mode DC-to-DC power converters are commonly used in the electronics industry to convert an available direct current (DC) level voltage to another DC level voltage. A switched mode converter provides a regulated DC output voltage to a load by selectively storing energy in an output inductor coupled to the load by switching the flow of current into the output inductor. A synchronous buck converter is a particular type of switched mode converter that uses two power switches, typically MOSFET transistors, to control the flow of current in the output inductor. A high-side switch selectively couples the inductor to a first power supply voltage while a low-side switch selectively couples the inductor to a second power supply voltage, such as ground. A filter capacitor coupled in parallel with the load reduces ripple of the output current. A pulse width modulation (PWM) control circuit is used to control the gating of the high-side and low-side switches in an alternating manner. Synchronous buck converters generally offer high efficiency and high power density, particularly when MOSFET devices are used due to their relatively low on-resistance. 
     For certain applications having demanding current requirements, it is known to combine plural synchronous buck converter modules together in multi-phase configurations operated in an interleaf mode. The output inductors of each of the buck converter modules are connected together to provide a single output voltage. The PWM control circuit provides a variable duty cycle control signal to each buck converter module in order to control its switching. The multiple modules are operated in a synchronous manner, with the respective high-side switches of each channel being switched on at different phases of a power cycle. Interleaf operation is advantageous in that it reduces the current ripple across the filter capacitor and makes the ripple frequency a multiple of the switching frequency, thereby enabling the use of smaller filter capacitors to reduce the ripple. Also, by spreading the output current among the multiple channels, the thermal load on the power semiconductor components of the power converter is reduced. 
     Recent advancements in microprocessors continue to drive a demand for power converters that supply increasingly low output voltages (e.g., less than 1.5 volts) at high load current (e.g., greater than 40 amps). To satisfy this demand, multi-phase power converters are operated at very high switching frequencies (e.g., greater than 100 kHz). But, at high switching frequencies, as the duty cycle is made very small (e.g., 10-40%), multi-phase power converters tend to exhibit poor ripple cancellation. Moreover, the high-side MOSFET devices have high switching losses due to the high switching voltage and current. To solve these problems, power converter topologies that utilize groups of coupled magnetic configurations have been proposed, such as the coupled buck converter and the tapped inductor buck converter. These topologies extend the power converter duty cycle, and have better ripple cancellation and lower switching losses due to lower switching current. 
     Nevertheless, these coupled magnetic topologies also have other drawbacks that make them less attractive as alternative designs. Coupled magnetic configurations are not functionally equivalent to buck converters, and operate analogously to flyback converter topologies in that the load current is partly supplied by the filter capacitor during the on-time of the power switches, hence requiring a larger filter capacitor. Further, coupled magnetic configurations have stability problems due to a right half plane zero that introduces an extra phase-shift of 90° into the control loop. Coupled magnetic configurations also have poor efficiency and large output ripple due to their discontinuous energy transfer to the output inductor. Lastly, coupled magnetic configurations use a single ended inductor in which the magnetic flux swing is unidirectional. This results in poor utilization of the transformer core since the core flux must reset naturally. In addition to these drawbacks, the relationship between the inductor turns ratio and duty cycle is not linear, so the duty cycle cannot be increased proportionally with the inductor turns ratio. 
     Accordingly, it would be desirable to provide a multi-phase power converter that can operate at high switching frequencies with effective ripple cancellation and reduced switching loss for high power density converter applications. It would also be desirable to provide such a multi-phase power converter having a simple, true buck-derived topology. 
     SUMMARY OF THE INVENTION 
     The present invention overcomes these drawbacks of the prior art by providing a multi-phase power converter that can operate at high switching frequencies with effective ripple cancellation and reduced switching loss for high power density converter applications. 
     In an embodiment of the invention, the multiple-phase power converter comprises a non-isolated, double-ended transformer having a plurality of windings, a high-side switch portion, a low-side switch portion, an output portion, and a controller. The high-side switch portion includes a first power switch connecting an input voltage source (V IN ) to a virtual phase node through a first winding of the plurality of windings and a second power switch connecting the input voltage source to the virtual phase node through a second winding of the plurality of windings. The first and second windings are arranged with opposite polarity. The low-side switch portion includes a third power switch connecting the virtual phase node to ground through a third winding of the plurality of windings and a fourth power switch connecting the virtual phase node to ground through a fourth winding of the plurality of windings. The third and fourth windings are arranged with opposite polarity. The output portion includes an output inductor connecting the virtual phase node to an output terminal providing an output voltage (V OUT ). 
     The controller is adapted to control operations of the first, second, third and fourth power switches such that the first and second power switches are enabled in respective alternating phases. The third switch is disabled when the second switch is enabled so that current flows concurrently through both the second and fourth windings to the output inductor during a first phase, and the fourth switch is disabled when the first switch is enabled so that current flows concurrently through the first and third windings to the output inductor. In a preferred embodiment, the first and second power switches are driven by respective control signals having a duty cycle of approximately 25% and a relative phase difference of 180°. Likewise, the third and fourth power switches are driven by respective control signals having a duty cycle of approximately 75% and a relative phase difference of 180°. 
     The multiple-phase power converter may further include a snubber circuit electrically connected to the high-side portion to reduce voltage ringing or spikes due to leakage inductance of the double-ended transformer. The snubber circuit may further comprise a low pass filter electrically connected to the first and second power switches. Alternatively, the snubber circuit may comprise a diode and a parallel-connected resistor and capacitor connected electrically between each of the first and second power switch or across each of the first and second winding. The multiple-phase power converter may further comprise a capacitor electrically connected between the first and second power switches to provide clamping of the high-side portion. 
     In another embodiment of the invention, the multiple-phase power converter is adapted for four-phase operation, and comprises a non-isolated, double-ended transformer having a plurality of windings, a first high-side switch portion, a first low-side switch portion, a first output portion, a second high-side switch portion, a second low-side switch portion, a second output portion and a controller. The first, second, fifth and sixth power switches are driven by respective control signals having a duty cycle of approximately 25% and a relative phase difference of 90°. The third, fourth, seventh and eighth power switches are driven by respective control signals having a duty cycle of approximately 75% and a relative phase difference of 90°. 
     A more complete understanding of the multi-phase power converter will be afforded to those skilled in the art, as well as a realization of additional advantages and objects thereof, by a consideration of the following detailed description of the preferred embodiment. Reference will be made to the appended sheets of drawings that will first be described briefly. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a schematic diagram of a two-phase DC-to-DC power converter in accordance with an embodiment of the invention; 
     FIG. 2 is a timing diagram illustrating switch control signals and other voltage and current waveforms of the power converter of FIG. 1; 
     FIGS. 3A-3C are schematic diagrams of alternative snubber circuits for use with the power converter of FIG. 1; 
     FIG. 4 is a schematic diagram of a two-phase DC-to-DC power converter having a clamping circuit in accordance with an alternative embodiment of the invention; 
     FIG. 5 is a timing diagram illustrating switch control signals for the power converter of FIG. 4; 
     FIG. 6 is a schematic diagram of a four-phase DC-to-DC power converter in accordance with an alternative embodiment of the invention; 
     FIG. 7 is a timing diagram illustrating switch control signals and other voltage and current waveforms of the power converter of FIG. 6; 
     FIG. 8 is a block diagram of a multi-module power supply system in accordance with an embodiment of the invention; and 
     FIG. 9 is a block diagram of a single module of the multi-module power supply system of FIG.  8 . 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     The present invention satisfies the need for a multi-phase power converter that can operate at high switching frequencies with effective ripple cancellation and reduced switching loss for high power density converter applications. In the detailed description that follows, like element numerals are used to describe like elements illustrated in one or more of the figures. 
     Referring first to FIG. 1, a two-phase DC-to-DC power converter  10  is illustrated in accordance with an embodiment of the invention. The DC-to-DC power converter  10  receives an input voltage (V IN ) and provides an output voltage (V OUT ) and output current (i O ) to a load (not shown). A capacitor  34  is electrically connected in parallel with the load to provide smoothing of the output voltage V OUT . The power converter  10  includes a pair of high-side power switches  22 ,  24  and a pair of low-side power switches  26 ,  28 . The high-side power switches  22 ,  24  and the low-side power switches  26 ,  28  are generally provided by MOSFET devices. A non-isolated, double-ended transformer is interposed with the high-side and low-side power switches, and includes interleaved windings  12 ,  14 ,  16 , and  18 . Windings  12 ,  14  have a turns ratio of N:1 with respect to windings  16 ,  18 . The drain terminals of the high-side power switches  22 ,  24  are electrically connected to the input voltage source V IN  through respective windings  12 ,  14 . The windings  12 ,  14  are arranged in parallel with opposite polarity. The source terminals of each of the high-side power switches  22 ,  24  are electrically connected to the drain terminals of the low-side power switches  26 ,  28  through respective windings  16 ,  18 . The windings  16 ,  18  are arranged in parallel with opposite polarity. The source terminals of the low-side power switches  26 ,  28  are electrically connected to ground. A power phase node (V P ) is defined at the junction between the sources of the high-side power switches  22 ,  24  and the windings  16 ,  18 . An output inductor  32  is connected in series between the power phase node and the load. 
     The gate terminals of the high-side and low-side power switches  22 ,  24 ,  26 ,  28  are connected to a control circuit (not shown), such as a pulse width modulator (PWM), that provides control signals to control the duty cycle applied to the power switches. The control signals are illustrated in FIG. 2, in which M 1  is the signal applied to high-side power switch  22 , M 3  is the signal applied to high-side power switch  24 , M 2  is the signal applied to low-side power switch  26 , and M 4  is the signal applied to low-side power switch  28 . Control signals M 1  and M 3  have a duty cycle of approximately 25% and have a relative phase difference of 180°. Conversely, control signals M 2  and M 4  have a duty cycle of approximately 75% and have a relative phase difference of 180°. FIG. 2 also illustrates other current and voltage measurements of the power converter  10  that vary in accordance with control signals, including the drain-to-source voltage across high-side power switch  22  (V ds M 1 ), the current from drain-to-source through high-side power switch  22  (i ds M 1 ), the drain-to-source voltage across low-side power switch  26  (V ds M 2 ), the current from drain-to-source through low-side power switch  26  (i ds M 2 ), and the voltage at the power phase node (V P ). 
     In a first phase of the power cycle, control signal M 1  goes high while control signal M 4  goes low, causing high-side power switch  22  to conduct and low-side power switch  28  to shut off. Control signal M 2  is already in a high state, so low-side power switch  26  is already conducting. This causes current to flow through winding  12  and high-side power switch  22  to the load, and through low-side power switch  26  and winding  16  to the load at the same time. The ratio of current flowing through these two paths is determined by the turns ratio of the double-ended transformer. At the end of this part of the power cycle, control signal M 1  goes low and control signal M 4  goes high. The stops the flow of current through winding  12  and high-side power switch  22 . Current continues to be delivered through windings  16  and  18  to the load as the circuit free-wheels to reset the transformer core. 
     Next, in a second phase of the power cycle, control signal M 3  goes high while control signal M 2  goes low, and a similar process occurs. Specifically, high-side power switch  24  conducts and low-side power switch  26  is shut off. Control signal M 4  is already in a high state, so low-side power switch  28  is already conducting. This causes current to flow through winding  14  and high-side power switch  24  to the load, and through low-side power switch  28  and winding  18  to the load at the same time. As before, the ratio of current flowing through these two paths is determined by the turns ratio of the double-ended transformer. At the end of this part of the power cycle, control signal M 3  goes low and control signal M 2  goes high. This stops the flow of current through winding  14  and high-side power switch  24 . Current continues to be delivered through windings  16  and  18  to the load as the circuit free-wheels to reset the transformer core. 
     It should be appreciated that the magnetic flux swing in the double-ended transformer is bi-directional. In other words, the transformer is being actively driven in two directions, as opposed to single-ended transformers that are driven in one direction and allowed to reset its core flux naturally. The double-ended transformer has the advantage of utilizing the core volume more efficiently, and therefore allowing the use of a physically smaller transformer core. The frequency of the voltage ripple across the output inductor  32  is twice that of the switching frequency of each individual phase. There is a linear relationship between the conversion ratio of the power converter and the duty cycle, as defined by the following equation: 
     
       
           V   OUT   /V   IN   =D /( N +1) 
       
     
     wherein D is the duty cycle of each phase, and N is the turns ratio of the non-isolated double ended transformer. Thus, the duty cycle can be increased proportionally with the transformer turns ratio. Moreover, the switching loss of the high-side power switches  22 ,  24  corresponds to N/(N+1) 2  that of the corresponding switching loss in a conventional buck converter. 
     There are many advantages of the present power converter topology. The topology achieves a true buck equivalent in that the output current is continuously delivered from the input side to the load. The double-ended transformer makes efficient use of the core volume. Substantial ripple cancellation is achieved, thereby reducing the output capacitor requirements. There is significantly reduced high-side power device switching loss, which enables the high frequency operation without thermal degradation to the high-side power devices. Lastly, the power converter topology provides fast transient response as compared with coupled inductor or flyback equivalent topologies. 
     In an embodiment of the invention, the power converter  10  may further include a snubber circuit. Even though the magnitude of the ripple on the output inductor  32  is reduced by half (since the frequency is doubled), it may still be desirable to reduce the ripple further. The ripple is due to leakage inductance of the transformer that keeps current flowing through the high-side power switch  22  after the switch has shut off, resulting in voltage ringing or spikes. A snubber circuit is known to reduce these undesirable effects. FIGS. 3A-3C illustrate various exemplary embodiments of snubber circuits used in association with the power converter  10 . In FIG. 3A, the snubber circuit includes a resistor  46  and capacitor  48  connected in series between the drain and source terminals of the high-side power switch  22 . The resistor.  46  and capacitor  48  provide a low-pass filter that dissipates the high frequency energy of the leakage inductance voltage spike. In FIG. 3B, the snubber circuit includes a diode  52  and parallel-connected resistor  54  and capacitor  56  connected in series between the drain and source terminals of the high-side power switch  22 . The snubber circuit serves to slow up the voltage rise time across the high-side power switch  22 . FIG. 3C shows a similar snubber circuit including a diode  62  and parallel-connected resistor  64  and capacitor  66  connected in parallel with transformer winding  12 . Other snubber circuits generally known in the art could also be advantageously utilized. 
     As known in the art, the control circuit typically receives feedback signals that are used to regulate the output voltage (V OUT ) and output current (i O ). The feedback signals may include a voltage error signal and a current sense signal. The voltage error signal corresponds to a difference between the output voltage (V OUT ) and a reference, and the current sense signal corresponds to the output current (i O ) being delivered to the load. It is anticipated that the power converter  10  include known current sensing techniques in order to generate the current sense signal, such as a sensing resistor included in series with the output inductor or in series with the drain of the high-side power switch. The current sense signal may be derived by monitoring the voltage drop across the sensing resistor. Alternatively, a filter including a resistor and capacitor may be disposed in parallel with the output inductor so that the instantaneous voltage across the capacitor can be made equal to the voltage across the DC resistance of the inductor and thereby proportional to the instantaneous current through the output inductor. It is also known to use the on-state resistance (R DSON ) between source and drain terminals of the high-side power switches as sensing resistors. 
     In a preferred embodiment of the invention, a current sensor includes a filter connected to the phase node that includes the on-state resistance of both the high-side and low-side power switches. In view of the increased resistance of the current sensor, the voltage of the current sense signal is increased and thereby provides a cleaner signal referenced to the output current that is less susceptible to noise than the aforementioned conventional current sense circuits. An example of the current sensor is provided by U.S. Pat. No. 6,441,597 for “Method And Apparatus For Sensing Output Inductor Current In A DC-to-DC Power Converter,” issued Aug. 27, 2002, which is incorporated by reference herein. The equivalent sense resistance of the current sensor can be expressed using the following equation: 
     
       
           R   EQUIV =( R   TOP /( N +1)) D +( R   BOT /2)(1 −D )+ RL   
       
     
     wherein R TOP  is the on-state resistance (R DSON ) of the high-side power switch plus the winding resistance of the winding in series with the high-side power switch, R BOT  is the on-state resistance (R DSON ) of the low-side power switch plus the winding resistance of the winding in series with the low-side power switch, RL is the output inductor series resistance, and N is the non-isolated transformer turns ratio. 
     FIG. 4 illustrates an alternative embodiment of a two-phase DC-to-DC power converter  100  that includes a clamped topology. As in the preceding embodiment, the power converter  100  includes a pair of high-side power switches  122 ,  124  and a pair of low-side power switches  126 ,  128 . A non-isolated, double-ended transformer is interposed with the high-side and low-side power switches, and includes interleaved windings  112 ,  114 ,  116 , and  118 . Windings  112 ,  114  have a turns ratio of N:1 with respect to windings  116 ,  118 . In this embodiment, the relative position of the high-side power switch  122  and winding  112  is reversed with respect to the previous embodiment. Specifically, the drain terminal of the high-side power switch  122  is directly connected to the input voltage source V IN  with the source terminal of high-side power switch electrically connected to the power phase node (V P ) through winding  112 , which is in turn connected to drain terminals of the low-side power switches  126 ,  128  through respective windings  116 ,  118 . The drain terminal of the high-side power switch  124  is connected to the input voltage source V IN  through the winding  114  in the same manner as the previous embodiment. The source terminal of high-side power switch  124  is electrically connected to the power phase node (V P ). A capacitor  136  is connected between the source terminal of the high-side power switch  122  and the drain terminal of the high-side power switch  124 . 
     As with the previous embodiment, the gate terminals of the high-side and low-side power switches  122 ,  124 ,  126 ,  128  are connected to a control circuit that provides control signals to control the duty cycle applied to the power switches. The control signals are illustrated in FIG. 5, in which M 1  is the signal applied to high-side power switch  122 , M 3  is the signal applied to high-side power switch  124 , M 2  is the signal applied to low-side power switch  126 , and M 4  is the signal applied to low-side power switch  128 . Control signals M 1  and M 3  have a duty cycle of approximately 25% and have a relative phase difference of 180°. Conversely, control signals M 2  and M 4  have a duty cycle of approximately 75% and have a relative phase difference of 180°. Since the drain terminal of the high-side power switch  122  is directly connected to the input voltage source V IN , it will be further necessary to drive the high-side power switch  122  at a higher voltage than the other power switches, such as using a bootstrap driver as is well known in the art. 
     The operation of the two-phase DC-to-DC power converter  100  is substantially the same as the previous embodiment, with the following difference. When the control signal M 1  goes low to shut off high-side power switch  122 , current continues to flow from the winding  112  through capacitor  136  and the body diode of high-side power switch  124 . Similarly, when the control signal M 3  goes low to shut off high-side power switch  124 , current continues to flow from the winding  114  through capacitor  136  and the winding  112 . The capacitor  136  therefore provides a clamping function by preventing the voltage at the power phase node (V P ) from ringing. 
     FIG. 6 illustrates a four-phase DC-to-DC power converter  200  in accordance with another embodiment of the invention. The DC-to-DC power converter  200  receives an input voltage (V IN ) and provides an output voltage (V OUT ) and output current (i O ) to a load (not shown). A capacitor  234  is electrically connected in parallel with the load to provide smoothing of the output voltage V OUT . The power converter  200  includes a first pair of high-side power switches  222 ,  224 , a first pair of low-side power switches  226 ,  228 , a second pair of high-side power switches  252 ,  254 , and a second pair of low-side power switches  256 ,  258 . Two non-isolated, double-ended transformers are interposed with the high-side and low-side power switches. The first transformer includes interleaved windings  212 ,  214 ,  216 , and  218 , and the second transformer includes windings  242 ,  244 ,  246  and  248 . The first transformer windings  212 ,  214  have a turns ratio of N:1 with respect to windings  216 ,  218 , and the second transformer windings  242 ,  244  have a turns ratio of N:1 with respect to windings  246 ,  248 . 
     The drain terminals of the first high-side power switches  222 ,  224  are electrically connected to the input voltage source V IN  through respective windings  212 ,  214 . The windings  212 ,  214  are arranged in parallel with opposite polarity. The source terminals of each of the first high-side power switches  222 ,  224  are electrically connected to the drain terminals of the first low-side power switches  226 ,  228  through respective windings  216 ,  218 . The windings  216 ,  218  are arranged in parallel with opposite polarity. The source terminals of the first low-side power switches  226 ,  228  are electrically connected to ground. A first power phase node (V P1 ) is defined at the junction between the sources of the first high-side power switches  222 ,  224  and the windings  216 ,  218 . A first output inductor  232  is connected in series between the first power phase node and the load. 
     Likewise, the drain terminals of the second high-side power switches  252 ,  254  are electrically connected to the input voltage source V IN  through respective windings  242 ,  244 . The windings  242 ,  244  are arranged in parallel with opposite polarity. The source terminals of each of the second high-side power switches  252 ,  254  are electrically connected to the drain terminals of the second low-side power switches  256 ,  258  through respective windings  246 ,  248 . The windings  246 ,  248  are arranged in parallel with opposite polarity. The source terminals of the second low-side power switches  256 ,  258  are electrically connected to ground. A second power phase node (V P2 ) is defined at the junction between the sources of the second high-side power switches  252 ,  254  and the windings  246 ,  248 . A second output inductor  236  is connected in series between the second power phase node and the load. 
     As described above, the gate terminals of the high-side and low-side power switches  222 ,  224 ,  226 ,  228 ,  252 ,  254 ,  256 ,  258  are connected to a control circuit that provides control signals to control the duty cycle applied to the power switches. The control signals are illustrated in FIG. 7, in which M 1  is the signal applied to high-side power switch  222 , M 3  is the signal applied to high-side power switch  224 , M 2  is the signal applied to low-side power switch  226 , M 4  is the signal applied to low-side power switch  228 , M 5  is the signal applied to high-side power switch  252 , M 7  is the signal applied to high-side power switch  254 , M 6  is the signal applied to low-side power switch  256 , and M 8  is the signal applied to low-side power switch  258 . Control signals M 1  through M 4  are substantially the same as described above with respect to the two-phase power converter. Control signals M 5  and M 7  have a duty cycle of approximately 25% and have a relative phase difference of 180°, and are shifted in phase with respect to control signals M 1  and M 3  by 90°, respectively. Conversely, control signals M 6  and M 8  have a duty cycle of approximately 75% and have a relative phase difference of 180°, and are shifted in phase with respect to control signals M 2  and M 4  by 90°, respectively. FIG. 7 also illustrates other current measurements of the power converter  200  that vary in accordance with control signals, including the current though first output inductor  232  (iL 1 ) and current through second output inductor  236  (iL 2 ). The output current (I O ) corresponds to the sum of iL 1  and iL 2 , also shown in FIG.  7 . 
     In a first phase of the power cycle, control signal M 1  goes high while control signal M 4  goes low, causing first high-side power switch  222  to conduct and first low-side power switch  228  to shut off. Control signal M 2  is already in a high state, so first low-side power switch  226  is already conducting. This causes current to flow through winding  212  and high-side power switch  222  to the load, and through low-side power switch  226  and winding  216  to the load at the same time. The ratio of current flowing through these two paths is determined by the turns ratio of the double-ended transformer. At the end of this part of the power cycle, control signal M 1  goes low and control signal M 4  goes high. This stops the flow of current through winding  212  and first high-side power switch  222 . Current continues to be delivered through windings  216 ,  218  to the load as the circuit free-wheels. 
     In a second phase of the power cycle, control signal M 5  goes high while control signal M 8  goes low, causing second high-side power switch  252  to conduct and second low-side power switch  258  to shut off. Control signal M 6  is already in a high state, so second low-side power switch  256  is already conducting. This causes current to flow through winding  242  and high-side power switch  252  to the load, and through low-side power switch  256  and winding  246  to the load at the same time. The ratio of current flowing through these two paths is determined by the turns ratio of the double-ended transformer. At the end of this part of the power cycle, control signal M 5  goes low and control signal M 8  goes high. This stops the flow of current through winding  242  and high-side power switch  252 . Current continues to be delivered through windings  246 ,  248  to the load as the circuit free-wheels. 
     In a third phase of the power cycle, control signal M 3  goes high while control signal M 2  goes low. First high-side power switch  224  conducts and first low-side power switch  226  is shut off. Control signal M 4  is already in a high state, so first low-side power switch  228  is already conducting. This causes current to flow through winding  214  and high-side power switch  224  to the load, and through low-side power switch  228  and winding  218  to the load at the same time. As before, the ratio of current flowing through these two paths is determined by the turns ratio of the double-ended transformer. At the end of this part of the power cycle, control signal M 3  goes low and control signal M 2  goes high. This stops the flow of current through winding  214  and high-side power switch  224 . Current continues to be delivered through windings  216 ,  218  to the load as the circuit free-wheels. 
     Lastly, in a fourth phase of the power cycle, control signal M 7  goes high while control signal M 6  goes low. Second high-side power switch  244  conducts and second low-side power switch  256  is shut off. Control signal M 8  is already in a high state, so second low-side power switch  258  is already conducting. This causes current to flow through winding  244  and high-side power switch  254  to the load, and through low-side power switch  258  and winding  248  to the load at the same time. As before, the ratio of current flowing through these two paths is determined by the turns ratio of the doubleended transformer. At the end of this part of the power cycle, control signal M 7  goes low and control signal M 6  goes high. This stops the flow of current through winding  244  and high-side power switch  254 . Current continues to be delivered through windings  246 ,  248  to the load as the circuit free-wheels. 
     With a per phase duty cycle of 25%, the four-phase power converter  200  provides true ripple cancellation and symmetrical transient response. It should be appreciated that the four-phase power converter  200  could further include a snubber circuit and/or current sense circuit as described above. 
     FIG. 8 illustrates the present topology utilized in a multi-module power supply system  300 . Modules  310 ,  320 ,  350  correspond to the two or four-phase power converters described above. Each module  310 ,  320 ,  350  receives a common input voltage (V IN ) and provides a respective output voltage (V OUT1 -V OUTn ) that is combined to supply a load (not shown). A current share bus also couples the modules  310 ,  320 ,  350 , and provides a current share signal used as a feedback control signal to manage the output current delivered to the load. 
     FIG. 9 illustrates one of the modules (e.g., module  310 ) of FIG. 8 in greater detail. The exemplary module  310  comprises a four-phase power converter as described above with respect to FIG. 6, although it should be appreciated that a two-phase or other n-phase power convert in accordance with the present invention could also be advantageously utilized. Module  310  further comprises a control integrated circuit (IC)  302  that provides timing and control signals to a plurality of MOSFET drivers  304 A- 304 D. The MOSFET drivers  304 A- 304 D in turn deliver control signals to respective power MOSFETs  306 A- 306 D. The power MOSFETs  306 A- 306 D correspond generally to the four pairs of power switches described above with respect to FIG.  6 . The power MOSFETs  306 A- 306 D are electrically connected with transformer/inductors  308 A,  308 B, which corresponds to the non-isolated, double-ended transformers and output inductors described above. Lastly, the module includes capacitor  312  electrically connected in parallel with the load to provide smoothing of the output voltage V OUT . 
     Having thus described a preferred embodiment of a multi-phase power converter that can operate at high switching frequencies with effective ripple cancellation and reduced switching loss for high power density converter applications, it should be apparent to those skilled in the art that certain advantages of the described method and apparatus have been achieved. It should also be appreciated that various modifications, adaptations, and alternative embodiments thereof may be made within the scope and spirit of the present invention. The invention is further defined by the following claims.