Abstract:
This invention provides apparatus and methods for causing a fluorescent lamp drive circuit to provide a continuous drive signal over a first (high) range of lamp intensity, and a pulse width modulated (PWM) drive signal over a second (low) range of lamp intensity, with a smooth transition between continuous and PWM drive that is unnoticeable to the user. This invention also provides fluorescent lamp circuits that include lamp intensity control circuitry, fluorescent lamp drive circuitry and a fluorescent lamp, the lamp intensity control circuitry providing control signals that cause the fluorescent lamp drive circuit to provide a continuous drive signal over a first (high) range of lamp intensity, and a PWM drive signal over a second (low) range of lamp intensity, with a smooth transition between continuous and PWM drive that is unnoticeable to the user.

Description:
BACKGROUND OF THE INVENTION 
     This invention relates to methods and apparatus for controlling the intensity of a fluorescent lamp. More particularly, this invention relates to methods and apparatus for providing control signals for a fluorescent lamp drive circuit to control the intensity of a fluorescent lamp. This invention also relates to fluorescent lamp circuits that include lamp intensity control circuitry, fluorescent lamp drive circuitry and a fluorescent lamp. 
     Fluorescent lamps increasingly are being used to provide efficient and broad-area visible light. For example, fluorescent lamps are used to back-light or side-light liquid crystal displays used in portable computer displays and flat panel liquid crystal displays. Fluorescent lamps also have been used to illuminate automobile dashboards and may be used with battery-driven, emergency-exit lighting systems. 
     Fluorescent lamps are useful in these and other low-voltage applications because they are more efficient, and emit light over a broader area, than incandescent lamps. Particularly in applications requiring long battery life, such as portable computers, the increased efficiency of fluorescent lamps translates into extended battery life, reduced battery weight, or both. 
     Liquid crystal computer displays typically are illuminated using a fluorescent lamp, such as a cold cathode fluorescent lamp (CCFL) that requires a high voltage, low current power source, and requires a much higher voltage to start than it does to maintain illumination. To insure a long lifetime, the lamp must not be operated above a maximum or below a minimum current. If a CCFL is operated at high current, the lamp becomes stressed and the lamp lifetime reduces. If a CCFL is operated at low current, the gaseous components inside the lamp will not fully ionize, and the lamp will slowly poison itself. In addition, at low currents, the lamp illumination tends to become uneven. Indeed, at low currents, the lamp may experience a so-called “thermometer effect,” in which one end of the lamp is dark. 
     Previously known fluorescent lamp drive circuits typically provide a continuous drive signal to illuminate a CCFL. To vary the intensity of a CCFL, the magnitude of the continuous drive current may be varied. Thus, to adjust the brightness of a liquid crystal computer display that includes a CCFL, the magnitude of the continuous drive current may be reduced to dim the display, or increased to brighten the display. Because of the lamp&#39;s narrow operating current range, however, a display that uses a CCFL has a narrow dimming range. 
     One previously known alternative to this continuous technique uses pulse width modulation (PWM) to extend the dimming range of a fluorescent lamp. That is, rather than varying the magnitude of a continuous drive signal to the lamp, the drive circuitry provides a drive signal that switches the lamp ON and OFF from maximum current to zero current at a fixed frequency. To control the lamp intensity, the drive circuit varies the duty cycle of the drive signal. Thus, a 100% duty cycle provides maximum bulb brightness, whereas a lower duty cycle effectively dims the lamp. PWM techniques extend the dimming range of the lamp without problems associated with uneven illumination at the low end of the dimming range. 
     To prevent noticeable flicker or interaction with ambient lighting, the PWM frequency must be approximately 100 to 200 Hz. A problem with this PWM technique is that except when the drive circuit operates the lamp at maximum brightness, the drive circuit always switches the lamp ON at maximum current and OFF at zero current at a 100 to 200 Hz rate. Constantly switching the lamp from OFF to ON requires that the drive circuitry repeatedly supply the high voltage necessary to start the lamp, which stresses the lamp and drive circuitry, and limits lamp lifetime. 
     In view of the foregoing, it would therefore be desirable to provide methods and apparatus for controlling the intensity of a fluorescent lamp without reducing the lamp&#39;s lifetime. 
     It further would be desirable to provide methods and apparatus that combine the advantages of the continuous and PWM techniques for controlling lamp intensity. 
     SUMMARY OF THE INVENTION 
     It is an object of this invention to provide methods and apparatus for controlling the intensity of a fluorescent lamp without reducing the lamp&#39;s lifetime. 
     It further is an object of this invention to provide methods and apparatus that combine the advantages of the continuous and PWM techniques for controlling lamp intensity. 
     These and other objects are accomplished in accordance with the principles of the present invention by providing control signals for a fluorescent lamp drive circuit. The control signals may be used to cause a fluorescent lamp drive circuit to provide a continuous drive signal over a first (high) range of lamp intensity, and a PWM drive signal over a second (low) range of lamp intensity, with a smooth transition between continuous and PWM drive that is unnoticeable to the user. 
     In addition, this invention provides fluorescent lamp circuits that include lamp intensity control circuitry, fluorescent lamp drive circuitry, a fluorescent lamp and current feedback circuitry, the lamp intensity control circuitry and current feedback circuitry providing control signals that cause the fluorescent lamp drive circuit to provide a continuous drive signal over a first (high) range of lamp intensity, and a PWM drive signal over a second (low) range of lamp intensity, with a smooth transition between continuous and PWM drive that is unnoticeable to the user. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The above and other objects and advantages of the present invention will be apparent upon consideration of the following detailed description, taken in conjunction with accompanying drawings, in which like reference characters refer to like parts throughout, and in which: 
     FIG. 1 is a block diagram of an exemplary lamp intensity control circuit that provides control signals in accordance with principles of the present invention; 
     FIG. 2 is a schematic diagram of a sawtooth waveform provided by the circuit of FIG. 1; 
     FIG. 3 is a current versus voltage transfer characteristic of the voltage-controlled current amplifier of FIG. 1; 
     FIG. 4 is a current versus voltage transfer characteristic of the circuit of FIG. 1; 
     FIGS. 5A and 5B are pulse width modulated currents of the circuit of FIG. 1; 
     FIG. 6 is a circuit diagram of an exemplary embodiment of a voltage-controlled current amplifier of the circuit of FIG. 1; 
     FIG. 7 is circuit diagram of an alternative exemplary embodiment of a voltage-controlled current amplifier of the circuit of FIG. 1; 
     FIG. 8 is a block diagram of a lamp circuit that includes the lamp intensity control circuit of FIG. 1; and 
     FIG. 9 is a schematic diagram of an exemplary embodiment of the lamp circuit of FIG.  8 . 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     This detailed description is organized as follows. First, an illustrative embodiment of a lamp intensity control circuit is described that provides control signals in accordance with this invention. Second, a fluorescent lamp circuit is described that includes a lamp intensity control circuit, fluorescent lamp drive circuit, fluorescent lamp and current feedback circuit in accordance with this invention. 
     FIG. 1 illustrates an embodiment of a lamp intensity control circuit for providing control signals of this invention. Control circuit  10  includes PWM generator  12 , comparator  14 , voltage-controlled current amplifier  16 , switches  18  and  20 , and inverter  22 . As described in more detail below, control circuit  10  also may include comparator  23 . Control circuit  10  receives input signals V PROG , V PWM , V MIN , I EXT  and I RMIN , and provides control signal I C  whose value is a function of V PROG . Control circuit  10  also may receive input signal V T  and may provide control signal VOFF whose value also is a function of V PROG . V PROG , V PWM , I EXT , I RMIN  and V T  are direct current (DC) signals. As described in more detail below, as a user adjusts the magnitude of V PROG , I C  varies to control the intensity of a fluorescent lamp. 
     PWM generator  12  has a first terminal coupled to V PWM  and a second terminal coupled to V MIN . As shown in FIG. 2, PWM generator  12  provides sawtooth output VPO that varies between V MIN  and V PWM . Alternatively, VPO may have a triangular waveform that varies between V MIN  and V PWM . VPO operates at a frequency f saw  that is sufficiently high that a controlled lamp has little noticeable flicker, but sufficiently low to permit a lamp drive circuit to settle when the drive circuit operates in PWM mode. Frequency f saw  preferably is between 100 to 200 Hz. 
     Referring again to FIG. 1, comparator  14  has a non-inverting input coupled to VPO, an inverting input coupled to V PROG , and an output VCOUT. Inverter  22  has an input coupled to VCOUT and provides output {overscore (VCOUT)}, which equals the complement of VCOUT. If V PROG  is greater than VPO, VCOUT is LOW and {overscore (VCOUT)} is HIGH. If V PROG  is less than VPO, VCOUT is HIGH and {overscore (VCOUT)} is LOW. VCOUT is coupled to switch  20 , and {overscore (VCOUT)} is coupled to switch  18 . 
     Voltage-controlled current amplifier  16  has input terminals coupled to I EXT , V PROG  and V PWM , and provides output current I 1  that varies as a function of V PROG , as shown in FIG.  3 . In particular, if V PROG  is greater than or equal to V MAX , I 1  equals I RMAX  (region  24  in FIG.  3 ). If V PROG  is less than V MAX  and greater than or equal to V PWM , I 1  varies linearly with V PROG  between a maximum value of I RMAX  and a clamp value I CLAMP  (region  26  in FIG.  3 ). In this region of operation, I 1  equals:                I   1     =       (       I   RMAX         V   MAX     -     V   MIN         )     ×     (       V   PROG     -     V   MIN       )               (   1   )                                
     When V PROG =V PWM , I 1 =I CLAMP . From equation (1), I CLAMP  equals:                I   CLAMP     =       (       I   RMAX         V   MAX     -     V   MIN         )     ×     (       V   PWM     -     V   MIN       )               (   2   )                                
     Finally, if V PROG  is less than V PWM , I 1  equals I CLAMP  (region  28  in FIG.  3 ). 
     Referring again to FIG. 1, signals VCOUT and {overscore (VCOUT)}, control switches  20  and  18  to switch currents I 1  and I RMIN  to provide control signal I C . Each of switches  18  and  20  may be any commonly used switch, such as a bipolar junction transistor (BJT), complementary metal oxide semiconductor (CMOS) transistor, or other suitable switch. As shown in FIG. 1, switch  18  is a BJT having a collector coupled to I 1 , a base coupled to {overscore (VCOUT)}, and an emitter coupled to I C . Switch  20  is a BJT having a collector coupled to I RMIN , a base coupled to VCOUT, and an emitter coupled to I C . 
     Control circuit  10  operates as follows. I RMAX  and I RMIN  set the maximum and minimum lamp current values, respectively, and V MIN  sets a lower limit for brightness adjustment. V PWM  may be selected in the range V MIN ≦V PWM ≦V MAX  to set clamp level I CLAMP  as shown in equation (2), above. 
     As shown in FIG. 4, as a user adjusts the magnitude of V PROG , I C  varies to set a desired lamp intensity. If V PROG  is greater than or equal to V MAX , V PROG  is greater than V PWM  and VPO, I 1  equals I RMAX , VCOUT is LOW, {overscore (VCOUT)} is HIGH, transistor  18  is ON, transistor  20  is OFF, and I C  equals the emitter current of transistor  18 , which substantially equals I RMAX  (region  30  in FIG.  4 ). 
     If V PROG  is less than V MAX  but greater than or equal to V PWM , V MAX  is greater than VPO, I 1  has a value that varies linearly with V PROG  between a maximum value of I RMAX  and a minimum value I CLAMP , VCOUT is LOW, {overscore (VCOUT)} is HIGH, transistor  18  is ON, transistor  20  is OFF, and I C  equals the emitter current of transistor  18 , which substantially equals I 1  (region  32  in FIG.  4 ). In this region of operation, control current I C  equals:                I   C     =       (       I   RMAX         V   MAX     -     V   MIN         )     ×     (       V   PROG     -     V   MIN       )               (   3   )                                
     If V PROG =V PWM , I C =I CLAMP . 
     If V PROG  is less or equal to V PWM  but greater than or equal to V MIN , VCOUT and {overscore (VCOUT)} are complementary PWM signals having a clock frequency of f saw  (and a period T saw =1/f saw ), transistors  20  and  18  switch ON and OFF as controlled by VCOUT and {overscore (VCOUT)}, and I C  is a PWM signal that switches between a maximum value of I CLAMP  and a minimum value of I RMIN , and has an average value shown as dashed region  34  in FIG.  4 . That is, I C  is a PWM signal that varies from 100% ON at V PROG =V PWM , to 100% OFF at V PROG =V MIN , and has an average value I C  shown by the dashed line in region  34 . Average value I C  equals:                  I   C     _     =         (         I   CLAMP     -     I   RMIN           V   PWM     -     V   MIN         )     ×     (       V   PROG     -     V   MIN       )       +     I   RMIN               (   4   )                                
     If V PROG =V PWM , (V PROG −V MIN ) equals (V PWM −V MIN ), and I C =I CLAMP . Thus, as V PROG  is reduced from just above V PWM  to just below V PWM , I C  smoothly transitions from region  32  to region  34  in FIG.  4 . 
     FIG. 5 illustrates I C  versus time for several values of V PROG  for V MIN ≦V PROG &lt;V PWM . As shown in FIG. 5A, if V PROG =V MIN +(0.7)×(V PWM −V MIN ), from equation (4), {overscore (I C +L )}=(0.7)×I CLAMP +(0.3)×I RMIN . As shown in FIG. 5B, if V PROG =V MIN +(0.1)×(V PWM −V MIN ), from equation (4), {overscore (I C +L )}=(0.1)×I CLAMP +(0.9)×I RMIN . 
     In PWM mode (region  34  in FIG.  4 ), control current I C  may be used to modulate the current of a fluorescent lamp between a maximum value of I CLAMP  and a minimum value of I RMIN . Because the lamp is not switched from fully OFF to fully ON, the lamp intensity may be controlled without overstressing the lamp. 
     Referring again to FIG. 1, if V PROG  is less than V MIN , VCOUT is HIGH, {overscore (VCOUT)} is LOW, transistor  18  is OFF, transistor  20  is ON, and I C  equals the emitter current of transistor  20 , which substantially equals I RMIN  (region  36  in FIG.  4 ). 
     Control circuit  10  also may include circuitry to provide a control signal that may be used to reduce lamp current to zero whenever V PROG  is below a predetermined value. For example, control circuit  10  may include comparator  23 , which has an inverting input coupled to V T , a non-inverting input coupled to V PROG , and an open-collector output VOFF. V T  is a threshold voltage chosen to set a value at which the lamp current should be reduced to zero, and typically is less than V MIN . If V PROG  is greater than V T , the output of the comparator is an open circuit. If V PROG  is less than V T , the output of the comparator is LOW. Alternatively, comparator  23  may be a conventional comparator having inputs coupled to V PROG  and V T  and providing an output signal that may be used to cause fluorescent lamp drive circuitry to shut OFF current to the fluorescent lamp whenever V PROG  is reduced below V T . 
     Referring to FIG. 6, an illustrative embodiment of voltage-controlled current amplifier  16  is described. Amplifier  16  includes first and second differential gain stages and a current-mirror output stage comprised of NPN transistors  40 ,  42 ,  48 ,  50 ,  60 ,  62 ,  64 ,  66 ,  72 ,  78  and  80 , PNP transistors  44  and  46 , resistors  52 ,  54 ,  88  and  92 , and current sources  56 ,  58 ,  68 ,  70 ,  74  and  76 . 
     The first differential amplifier includes transistors  40 ,  42 ,  44 ,  46 ,  48  and  50 , resistors  52  and  54 , and current sources  56  and  58 . The first differential amplifier has a first input at a base of transistor  48 , a second input at a base of transistor  50 , external current source I EXT  coupled to emitters of transistors  40  and  42 , and an output at a base of transistor  44 . In this exemplary embodiment, I EXT  conducts current I RMAX . Diode-connected transistors  44  and  46  and emitter degeneration resistors  52  and  54  serve as loads. Current sources  56  and  58  each conduct current I B1  whose value is chosen to keep emitter-follower transistors  48  and  50  biased ON. 
     The second differential amplifier includes transistors  60 ,  62 ,  64 ,  66 ,  72 ,  78  and  80 , resistor  92 , and current sources  68 ,  70 ,  74  and  76 . The second differential amplifier has a first input V BIAS  coupled to a base of transistor  72 , a second input V PROG  coupled to a base of transistor  78 , a third input V PWM  coupled to a base of transistor  80 , a first output at a collector of transistor  64  coupled to the first input of the first differential amplifier, and a second output at a collector of transistor  66  coupled to the second input of the first differential amplifier. 
     The output stage includes transistor  90  and resistor  88 , and has an input at a base of transistor  90  coupled to the output of the first differential amplifier, and an output at terminal I 1 . Transistor  90  and transistor  44  form a current mirror, and emitter degeneration resistors  52 ,  54  and  88  each have a value R 1  chosen to reduce the effect of any base-emitter voltage (V BE ) mismatch between transistors  44 ,  46  and  90 . 
     Resistor  92  has a value R 2 , current sources  74  and  76  conduct current I B1 , and current sources  68  and  70  conduct current I B2 . V BIAS  is a voltage source having a value of approximately (V MAX −V MIN )/2 (FIG.  4 ). Resistance R 2  and bias current I B2  have values selected so that the second differential amplifier has a linear range of operation that extends from approximately V MIN  to V MAX  (FIG.  4 ). 
     Amplifier  16  operates as follows. V MAX  has a value approximately equal to (V BIAS  +R 2 ×I B2 ). If V PROG  is greater than V MAX , transistors  64  and  80  are OFF, transistors  78  and  66  are ON, transistor  42  is OFF, transistors  40  and  48  are ON, and transistors  40  and  44  conduct current  10  substantially equal to current I EXT =I RMAX . Transistors  44  and  90  have substantially the same base-emitter area, and resistors  52  and  88  have substantially the same resistance R 1 . The base-emitter voltage of transistor  44  substantially equals the base-emitter voltage of transistor  90 , and therefore, I 1  substantially equals I RMAX . This corresponds to region  24  in FIG.  3 . 
     As V PROG  is reduced below V MAX , the voltages at the emitters of transistors  66  and  78  reduce, transistor  80  remains OFF, transistor  64  begins to conduct, and the second differential amplifier enters its linear range of operation. As a result, transistor  42  begins to conduct, and steers a portion of I EXT  away from transistors  40  and  44 . As a result, I 0  and I 1  reduce linearly with V PROG . This corresponds to region  26  in FIG.  3 . 
     As V PROG  is further reduced, the voltage at the base of transistor  78  approaches V PWM , and transistors  78  and  80  both conduct current. I 0  and I 1  continue to reduce with reductions in V PROG , until V PROG  is slightly less than V PWM . At that point, transistor  78  is OFF, and any further reductions in V PROG  produce no further reductions in I 0  or I 1 . V PWM  thus sets clamp level I CLAMP  for amplifier  16 . This corresponds to region  28  in FIG.  3 . 
     In this embodiment, resistor  88  and transistor  90  are rationed to resistor  52  and transistor  44  so that I 1 =I 0 . By modifying the ratios, I 1  may be made substantially equal to a multiple of 
     FIG. 7 shows an alternative embodiment of a voltage-controlled current amplifier in accordance with this invention that consumes less power than amplifier  16 , and provides a more accurate output current at maximum current levels. In particular, amplifier  16 ′ is similar to amplifier  16 , but resistor  88 ′ and transistor  90 ′ are rationed so that I 1 =5×I 0 . That is, transistor  90 ′ has a base-emitter junction area five times the size of the base-emitter junction area of transistors  44  and  46 , and resistor  881  has a resistance R 3  that is one-fifth the size of resistance R 1  (i.e., R 3 =R 1 /5). Further, to provide a maximum current I 1 =I RMAX , I EXT =I RMAX /5. Thus, the differential pair comprising transistors  40 ,  42 ,  44  and  46 , and resistors  52  and  54  operate at a lower current than in amplifier  16 . 
     Because transistor  40  operates at a lower current than in amplifier  16 , the collector current of transistor  40  may not by itself be sufficient to drive the base of transistor  90 ′. Thus, an amplifier including resistor  82 , transistor  84  and capacitor  86  is included to supply additional base drive for transistor  90 ′. Resistor  82  biases transistor  84  at a small current, and has a resistance R 4  that is much larger than R 1  and R 3  (e.g., R 4 =25×R 1 ). Capacitor  84  has a capacitance C to compensate the base-drive amplifier. 
     FIG. 8 illustrates an exemplary embodiment of a fluorescent lamp circuit that includes a lamp intensity control circuit in accordance with this invention. Circuit  100  includes control circuit  10 , low voltage DC source  110 , regulator  112 , high voltage inverter  114 , lamp  116 , current feedback circuit  118 , summing node  120 , and current-to-voltage converter  122 . 
     Low-voltage DC source  110  provides power for circuit  100 , and may be any source of DC power. For example, in the case of a portable computer such as a lap-top or notebook computer, DC source  110  may be one or more nickel-cadmium or nickel-hydride batteries providing 3-20 volts. Alternatively, if lamp circuit  100  is used with an automobile dashboard, DC source  110  may be a 12-14 volt automobile battery and power supply. 
     DC source  110  supplies low-voltage DC to regulator  112  and may provide low-voltage DC to inverter  114 . Regulator  112  may include any of a number of commercially available linear or switching regulators. As shown in FIG. 8, voltage regulator  112  includes switching regulator  124  and inductor  126 . Switching regulator  124  may be, for example, the LT-1072 switching regulator manufactured by Linear Technology Corporation, Milpitas, Calif., or other suitable switching regulator. When implemented using the LT-1072, switching regulator  124  includes feedback terminal FB adapted to receive a feedback signal by which the output of voltage regulator  112  can be controlled, and control terminal V C , by which the switching regulator may be placed in shutdown mode. 
     Voltage regulator  112  provides regulated low-voltage DC output I dc  to inverter  114 . Inverter  114  converts I dc  to a high-voltage, high-frequency AC output V AC  of sufficient magnitude to drive fluorescent lamp  116 . Fluorescent lamp  116  may be any type of fluorescent lamp. For example, in the case of lighting a display in a portable computer, fluorescent lamp  116  may be a cold- or hot-cathode fluorescent lamp. 
     Current feedback circuit  118  generates a feedback current I FB  that is proportional to fluorescent lamp current I L . Summing node  120  provides an error signal I E  proportional to the difference between control current I C  and feedback current I FB . Current-to-voltage converter  122  converts error signal I E  to voltage V FB , which is coupled to terminal FB of switching regulator  124 . This feedback loop causes the magnitude of lamp current I L  to be proportional to the control current I C , so that I E  is substantially zero. 
     FIG. 9 shows a schematic diagram of an exemplary embodiment of lamp circuit  100  of FIG.  8 . Switching regulator  124  is implemented using an LT-1072 switching regulator, although any other suitable switching regulator may be used. As shown in FIG. 9, switching regulator  124  includes pin V IN  coupled to low voltage DC source  110 , terminals E 1 , E 2  and GND coupled to GROUND, control terminal V C  coupled to open-collector output VOFF from lamp intensity control circuit  10  and coupled through capacitor  156  to GROUND, switched output pin V SW  coupled to inductor  126  and Schottky diode  154 , and feedback pin FB coupled to terminal I C  of lamp intensity control circuit  10  and capacitor  152 . 
     Inverter circuit  114  is a current-driven, high-voltage, push-pull inverter which converts DC power from low voltage DC source  110  to high-voltage, sinusoidal AC. Inverter circuit  114  is a self-oscillating circuit, and includes transistors  132  and  134 , capacitors  136  and  138 , and transformer  140 . Transistors  132  and  134  conduct out of phase and switch each time transformer  140  saturates. During a complete cycle, the magnetic flux density in the core of transformer  140  varies between a saturation value in one direction and a saturation value in the opposite direction. During the cycle time when the magnetic flux density varies from negative minimum to positive maximum, one of transistors  132  and  134  is ON. During the rest of the cycle time (i.e., when the magnetic flux density varies from positive maximum to negative minimum), the other transistor is ON. 
     Switching of transistors  132  and  134  is initiated when the magnetic flux density in transformer  140  begins to saturate. At that time, the inductance of transformer  140  decreases rapidly toward zero, with the result that a quickly rising high collector current flows in the transistor that is ON. This current spike is picked up by transformer bias winding  140   b  of transformer  140 . Because the base terminals of transistors  132  and  134  are coupled to bias winding  140   b  of transformer  140 , the current spike is fed back into the base of the transistor that produced the spike. As a result, that transistor drops out of saturation and into cutoff, and the transistor is turned OFF. Accordingly, the current in transformer  140  abruptly drops, and the transformer winding voltages then reverse polarity resulting in the turning ON of the other transistor that previously had been OFF. The switching operation is then repeated for this second transistor. 
     Transistors  132  and  134  alternately switch ON and OFF at a duty cycle of approximately 50 percent. Capacitor  136 , coupled between the collectors of transistors  132  and  134 , causes what would otherwise be square-wave-like voltage oscillation at the collectors of transistors  132  and  134  to be substantially sinusoidal. Capacitor  136 , therefore, operates to reduce radio-frequency (RF) emissions from the circuit. The characteristics of transformer  140 , capacitor  136 , fluorescent lamp  116 , and ballast capacitor  146  coupled to secondary winding  140   d  of transformer  140  primarily determine the frequency of oscillation. Capacitor  138  reduces the high frequency impedance so that transformer center tap  140   a  sees zero impedance at all frequencies. 
     Transformer  140  steps-up the sinusoidal voltage at the collectors of transistors  132  and  134  to produce at secondary winding  140   d  an AC waveform of sufficiently high voltage to drive fluorescent lamp  116  (shown coupled to secondary winding  140   d  through ballast capacitor  146 ). Ballast capacitor  146  inserts a controlled impedance in series with lamp  116  to minimize sensitivity of the circuit to lamp characteristics and to minimize exposure of fluorescent lamp  116  to DC components. 
     Inverter  114  and current-mode switching regulator circuit  124  thus operate to deliver a controlled AC current at high voltage to fluorescent lamp  116 . Inductor  126 , coupled between V SW  of regulator  124  and the emitters of transistors  132  and  134 , is an energy storage element for switching regulator circuit  124 . Inductor  126  also sets the magnitude of the collector currents of transistors  132  and  134  and, hence, the energy through primary winding  140   c  of transformer  140  that is delivered to lamp  116  via secondary winding  140   d . Schottky diode  154 , coupled between low voltage DC power source  110  and switched output pin V SW , maintains current flow through inductor  126  during the OFF cycles of switching regulator circuit  124 . Resistor  130  DC-biases the respective bases of transistors  132  and  134 . 
     Inverter  114  may be implemented using circuitry other than that illustrated in FIG. 9, For example, inverter  114  may be implemented using ceramic step-up transformer technologies. 
     Current feedback circuit  118  may be implemented in integrated circuit technology, and includes diode-connected transistor  148 , transistor  150  and diode-connected transistor  158 . Transistor  148  has its base and collector coupled to GROUND, and has its emitter coupled to lamp  116 . Transistor  150  has its collector coupled to summing node  120 , its base coupled to the base of transistor  148 , and its emitter coupled to lamp  116  and the emitter of transistor  148 . Transistor  158  has its base and collector coupled together and to lamp  116 , and its emitter coupled to GROUND. 
     Diode-connected transistor  148  and diode-connected transistor  158  half-wave rectify lamp current I L . Transistor  158  shunts positive portions of each cycle of I L  to GROUND, and transistor  148  shunts a fraction of negative portions of I L  to GROUND. In particular, transistor  148  and  150  form a current mirror, with the collector of transistor  150  conducting a fraction of the current conducted by the collector of transistor  148 . As shown in FIG. 9, the base-emitter area of transistor  148  is ten times the size of the base-emitter area of transistor  150 , and therefore the collector current of transistor  150  is approximately one-tenth the collector current of transistor  148 . As a result, feedback current I FB  equals the negative portions of I L , reduced in magnitude by approximately one-eleventh. 
     Error current I E  equals the difference between control current I C  and feedback current I FB . Current-to-voltage converter  122  comprises capacitor  152 , which provides voltage V FB  equal to the integral of error current I E . V FB  therefore is proportional to error current I E , and is coupled to feedback pin FB of switching regulator  125 . The above connections close the feedback control loop that regulates lamp current I L  to control the intensity of lamp  116 . 
     Upon start-up of circuit  100  of FIG. 9, voltage V FB  on feedback pin FB generally is below the internal reference voltage of regulator circuit  124  (i.e., 1.23 volts for the LT-1072 discussed above). Thus, full duty cycle modulation at the switched output pin V sw  of regulator circuit  124  occurs. As a result, transistors  132  and  134  and inductor  126  conduct current from center tap  140   a  of transformer  140 . This current is conducted in switched fashion to GROUND by the action of switching regulator  124 . This switching action controls lamp current I L , which is set by the magnitude of the feedback signal V FB  at the feedback terminal FB of switching regulator  124 . The feedback loop forces switching regulator  124  to modulate the output of inverter  114  to whatever value is required so that error current I E  is substantially zero. 
     The circuit of FIG. 9 may be implemented using commercially available components. For example, the circuit can be constructed and operated using the components and values set forth below: 
     
       
         
               
               
               
             
           
               
                   
                   
               
               
                   
                 Component 
                 Source or Value 
               
               
                   
                   
               
             
             
               
                   
                 Regulator 124 
                 LT-1072 
               
               
                   
                 Inductor 126 
                 300 μH (COILTRONICS CTX300-4) 
               
               
                   
                 Resistor 130 
                 1 KΩ 
               
               
                   
                 Transistors 132 &amp; 134 
                 MPS650 
               
               
                   
                 Capacitor 136 
                 low loss 0.02 microfarad 
               
               
                   
                   
                 (Metalized polycarb WIMA-FKP2 
               
               
                   
                   
                 (Germany) preferred) 
               
               
                   
                 Capacitor 138 
                 10 μF 
               
               
                   
                 Transformer 140 
                 SUMIDA-6345-020 (available 
               
               
                   
                   
                 from SUMIDA ELECTRIC (USA) 
               
               
                   
                   
                 CO., LTD., of Arlington 
               
               
                   
                   
                 Heights, Illinois) or 
               
               
                   
                   
                 COILTRONICS CTX110092-1 
               
               
                   
                   
                 (available from Coiltronics Incorporated, 
               
               
                   
                   
                 of Pompano Beach, Florida) 
               
               
                   
                 Capacitor 146 
                 33 pF, rated up to 3 KV 
               
               
                   
                 Transistor 148 
                 10X 
               
               
                   
                 Transistor 150 
                  1X 
               
               
                   
                 Schottky diode 154 
                 1N5818 
               
               
                   
                 Capacitor 156 
                 0.1 μF 
               
               
                   
                 Transistor 158 
                  1X 
               
               
                   
                   
               
             
          
         
       
     
     The above circuit components and values are merely illustrative. Other circuit components and values also may be used. 
     Persons of ordinary skill in the art will recognize that lamp intensity control circuits of this invention may be implemented using integrated circuit technology along with other circuitry. For example, a lamp intensity control circuit may be combined along with a regulator circuit, such as a current-mode switching regulator circuit, and a current feedback circuit on a single integrated circuit to provide a fluorescent lamp controller. 
     In addition, persons of ordinary skill in the art will recognize that lamp intensity control circuits and lamp circuits of the present invention can be implemented using circuit configurations other than those shown and discussed above. All such modifications are within the scope of the present invention, which is limited only by the claims that follow.