Abstract:
A current sensor includes coupled inductors that generate an output current responsive to a detected current. The coupled inductor is implemented in an integrated circuited. An integrator circuit generates a sensed voltage responsive to the output current.

Description:
TECHNICAL FIELD OF THE INVENTION  
       [0001]     The present invention relates to current sensors, and more particularly, to an improved current sensor having low loss, small size, low cost and high accuracy.  
       BACKGROUND OF THE INVENTION  
       [0002]     Within various circuit implementations, such as power supplies, there is often the need to detect a current provided at a particular point within a circuit to use as feedback for controlling other parts of the circuit. Various solutions are presently used to sense currents within electronic circuits but each of these suffer from various shortcomings. A first approach, illustrated in  FIG. 1 , utilizes a resistor  102  connected across the inputs of an operational amplifier  104  to provide a voltage V SENSE  that may be used to determine a current  106 . A low value resistor  102  may be used in the range of 10 milliohms. The problem with this approach is the high loss provided by the circuit. This may be overcome by reducing the resistor  102  to reduce the loss, however, this also reduces the signal V SENSE  that may be detected. The resistor  102  is not integrated and this type of circuit may be used to sense current within direct current (DC) applications.  
         [0003]     Referring now to  FIG. 2 , there is illustrated a further prior art system utilizing a hall effect device  202  connected across the inputs of an operational amplifier  204 . The hall effect device  202  generates a voltage across the inputs of the operational amplifier  204  responsive to the current  206  to provide the output signal V SENSE . While this approach has a low loss, the use of the hall effect device  202  causes the circuit to have a higher cost, and the accuracy and noise issues are greater within the hall device as the hall voltage is a small value. This circuit may also be used to detect current in a direct current (DC) system.  
         [0004]     Referring now to  FIG. 3 , there is illustrated the use of a magneto resistive sensor. The magneto resistive sensor consists of a magneto resistive element  302  connected across the inputs of operational amplifier  304  to detect the current  306 . The magneto resistive element  302  has the property that the resistance of the element changes with respect to the magnetic field caused by the current  306 . This circuit requires the use of special technology which raises the cost of the device. Additionally, accuracy issues arise even though the current may be sensed with very low loss.  
         [0005]     Referring now to  FIG. 4 , an alternative prior art method for detecting current is through the use of a current transformer  402  is illustrated. The current transformer has a primary side  404  with a single loop and a secondary side  406  with multiple loops. A load resistance  408  is in parallel with the secondary side  406  of the transformer  402 . The current transformer  402  is used to detect the current  410 . The transformer  402  creates an output current equal to Ip/n with Ip being the detected current and n being the turns ratio of the transformer  402 . The resistance of the transformer is reflected to the primary side with the ratio 1/n 2 . A current transformer will only work within alternating current (AC) circuits. While current transformers work well for detecting currents, they are large and have a medium loss level associated therewith. Thus, some method for detecting a current within a power electronic circuit that overcome the shortcomings of these prior art methods would be greatly desirable.  
         [0006]     Another method for measuring currents involves the use of a Rogowski coil. The voltage induced in a Rogowski coil is very small and easily disturbed when measured current is less than, for example, 100 Amps. A Rogowski current transducer has a number of advantages over the current transformer illustrated in  FIG. 4 . For example, it is linear, has no core saturation effects and has wide band width, wide measurement range and a simple structure. This Rogowski coil comprises a toroidal winding placed around a conductor being measured. It consists of a wire wound on a non-magnetic core. The coil is effectively a mutual inductor coupled to the inductor being measured where the output from the winding is an EMF proportional to the rate of change of current.  
       SUMMARY OF THE INVENTION  
       [0007]     The present invention disclosed and claimed herein, in one aspect thereof, comprises a current sensor. The current sensor includes coupled inductors for generating an output current responsive to a detected current. The coupled inductor is implemented in an integrated circuited. An integrator circuit generates a sensed voltage responsive to the output current. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0008]     For a more complete understanding of the present invention and the advantages thereof, reference is now made to the following description taken in conjunction with the accompanying Drawings in which:  
         [0009]      FIG. 1  illustrates a prior art current sensor;  
         [0010]      FIG. 2  illustrates a further prior art current sensor;  
         [0011]      FIG. 3  illustrates yet another prior art current sensor;  
         [0012]      FIG. 4  illustrates a further prior art current sensor;  
         [0013]      FIG. 5   a  illustrates a coil in close proximity with a large current carrying wire according to the present disclosure;  
         [0014]      FIG. 5   b  illustrates a perspective cut away view of the integrated circuit, including a coupled coil and wire;  
         [0015]      FIG. 6  is a cross-sectional view of a first embodiment an integrated current sensor package;  
         [0016]      FIG. 7  is a top view of the first embodiment of the integrated current sensor package;  
         [0017]      FIG. 8  is a simulation of the integrated current sensor illustrated in  FIGS. 6 and 7 ;  
         [0018]      FIG. 9  is a cross-sectional view of an alternative embodiment of the integrated current sensor package;  
         [0019]      FIG. 10  is a top view of the alternative embodiment of the integrated current sensor package of  FIG. 9 ;  
         [0020]      FIG. 11  is a simulation of the alternative embodiment of the integrated current sensor package illustrated in  FIGS. 9 and 10 ;  
         [0021]      FIG. 12  is a schematic diagram of the integrated current sensor;  
         [0022]      FIG. 13  is a schematic diagram illustrating the integrated current sensor within a switched power supply circuit;  
         [0023]      FIG. 14  is a timing diagram illustrating operation of the switched power supply circuit of  FIG. 13 ;  
         [0024]      FIG. 15  illustrates a further method for controlling the reset switch of the integrated current sensor;  
         [0025]      FIG. 16  is a top view of the further embodiment of the integrated current sensor package;  
         [0026]      FIG. 17  is a cross-sectional view of the embodiment of the integrated current sensor package in  FIG. 16 ; and  
         [0027]      FIG. 18  is a further cross-sectional view of the embodiment of the integrated current sensor package in  FIG. 16 .  
     
    
     DETAILED DESCRIPTION OF THE INVENTION  
       [0028]     Referring now to the drawings, and more particularly to  FIG. 5   a,  there is illustrated a coil  502  in close proximity with a large current carrying wire  504  such that the coil  502  and current carrying wire  504  act as coupled inductors. The coupled inductors, along with on-chip electronics which will be discussed herein below, allow the creation of the V SENSE  signal which is proportional to the input current I p  in a manner that has very low loss, is very small and is a low cost implementation. This provides a better solution than all of the implementations described with respect to  FIGS. 1-4 . The current provided through the current carrying wire may be up to 10 amps. The coil  502  is placed very near the current carrying wire  504  in order to create the inductive coupling between the wire  504  and coil  502 . The wire  504  only overlaps only one side of the coil  502  such that the winding are all going the same way and the magnetic flux will add together. This causes an induced current in the other side of the coil  502  that is not overlapped by the wire  504 .  
         [0029]     Referring now to  FIG. 5   b,  there is illustrated a perspective cut away view of the coil  502  and wire  504  illustrated in  FIG. 5   a.  In this configuration one of the coupled inductors is placed within a silicon dioxide layer  604  on top of a die  606  of an integrated circuit chip. The coil  502  consists of metal runs within the M5 layer of the silicon dioxide layer  604 . The wire  504  would rest on the silicon dioxide layer  604  in close enough proximity to the coil  502  such that the current passing through wire  504  would induce another current within the portion of the coil  502  over which the wire  504  was not located.  
         [0030]     There are multiple ways for implementing the coupled inductor configuration illustrated in  FIG. 1  within a chip package. The first of these comprises an on-chip solution with bumping copper as illustrated in  FIG. 6 . Flip chip bump houses can deposit a copper wire  602  on top of a silicon dioxide layer  604  of a die  606 . The copper wire  602  may comprise 15 μm of copper. The coil  502  is embedded within the M 5  layer of the silicon dioxide layer  604 .  
         [0031]     Referring now to  FIG. 7 , there is illustrated a top view of the package configuration. The copper wire  602  is placed upon the silicon dioxide layer  604  of the die  606  (not shown). The coil  502  is placed within the M 5  layer of the silicon dioxide  604  parallel to the wire  502 . Bond wires  702  connect the copper wire  502  on the die  606  with external outputs. The bond wires  702  support a maximum current of 1-2 amps, thus many bond wires are required to be connected to the copper wire  602  for higher currents. Additional bond wires  704  connect the silicon dioxide layer  604  of the die  606  to external output pins  706  of the chip. Using the above-described package configuration, a 10 amp sensor may be constructed.  
         [0032]     Referring now to  FIG. 8 , there is provided a simulation of the inductive coil package illustrated in  FIGS. 6 and 7 . The coil  802  on the primary side comprises a 500 pH coil. The coil  804  on the secondary side comprises a 2 μH coil. Connected to a first side of the 500 pH coil  802  is a 1.5 milli-ohm resistor in series with a 0.5 milli-ohm resistor  810  which comprises the parasitic capacitance of the coil  802 . Connected to one output side of the 2 μH coil  804  is a 20 kiliohm resistor  812  which comprises the parasitic capacitance of the coil  804 . The 0.5 milli-ohm transistor  810  comprises the resistance provided by the bond wires and package. Since the copper wire  602  is not too thick and lies very close to the coil  502  of the chip, coupling coefficients between the copper wire  602  and the coil  502  are very good, assuming there is a distance of approximately two micrometers from the M5 layer to the copper wire  602 .  
         [0033]     Referring now to  FIG. 9 , there is illustrated an alternative configuration wherein a package lead frame and flip chip configuration are used. A custom package lead frame may be designed as follows. The die  902  is placed upside down with the silicon dioxide layer  904  suspended a short distance above a large copper slug  906 . The copper slug  906  may have a large cross sectional area for low loss. In this embodiment the slug  906  has a 200×200 μm cross section. The die  902  is suspended above the copper slug  906  on solder balls  908 . The solder balls  908  rest on top of a lead frame  910 . When heat is applied to the circuit, the solder bumps  908  reflow causing the silicon dioxide layer  904  to rest directly upon the copper slug  906 . In this design, the die chip  902  would be bumped and then flipped.  
         [0034]     Referring now to  FIG. 10 , there is illustrated a top view with the silicon dioxide layer  904  resting on top of the copper slug  906 . Bond wires  1002  may then be connected to the silicon dioxide layer  904 . This design has a very low resistance.  
         [0035]     Referring now to  FIG. 11 , there is illustrated the simulation of the embodiment illustrated in  FIGS. 9 and 10 . In this simulation wherein a 200×200 μm copper slug  906  is utilized that is 3 μm away from the coil  502 , the primary side consists of a 520 pH coil  1102  in series with a 0.5 milli-ohm resistor  1104 . The secondary side consists of 2 μH coil  1106  in series with a 20 kili-ohm resistor  1108 . The coupling coefficient is reduced due to the lower current density in the slug.  
         [0036]     Referring now to  FIG. 16  there is illustrated a bottom view of a further configuration wherein a lead on chip configuration is used. The lead frame  1602  is connected to the die  1604  by bond wires  1606 . The wire  1608  in connected to the die  1604  by tape. The wire  1608 , a current carrying conductor, is coupled to a coil in the die  1604 .  
         [0037]     Referring now to  FIG. 17 , there is illustrated a cross sectional view of  FIG. 16  along line B-B. The die  1604  is connected to the wire  1608  via the tape  1702  as described previously. The lead frame  1602  connects to the die  1604  via bond wires  1606 . The tape  1702  is approximately 75 μm thick. The entire structure is contained within a mold compound  1704 . Referring now to  FIG. 18 , there is illustrated a cross sectional view of  FIG. 16  along line A-A.  
         [0038]     Referring now to  FIG. 12 , there is illustrated a schematic diagram of the electronic circuit necessary for recreating the V SENSE  signal when detecting the current I p  using the coupled indicator as illustrated in  FIG. 5 . The coupled inductor  1202  comprises either of the configuration packages described hereinabove or, alternatively, may comprise a different undescribed configuration package that places the coil in close proximity with the wire to inductively couple them together. The primary side is modeled by inductor  1204  in series with resistor  1206 . The secondary side is modeled by inductor  1208  which is connected to a resistor  1210 . The resistor  1210  is then connected to ground. The other side of inductor  1208  outputs the induced current I n  which is connected to the negative input of an operational amplifier  1212 . The positive input of operational amplifier  1212  is connected to ground.  
         [0039]     The current through the secondary is dominated by the resistive loss of resistor  1210  and is the derivative of the primary current. An integrator circuit  1218  is used to integrate the induced current I n . The integrator circuit  1218  consists of the operational amplifier  1212 , a capacitor  1214  connected between the output of operational amplifier  1212  and the negative input of operational amplifier  1212  and a reset switch  1216  connected between the output of operational amplifier  1212  and the negative input of operational amplifier  1212  in parallel with capacitor  1214 . Thus, the current I n  may be determined according to the equation:  
         I   n     =       Lm     R   1       ⁢       ⅆ   ip       ⅆ   t             
 
 By integrating on the capacitor  1214  an output voltage VSENSE is attained according to the following equation:  
         V   SENSE     =         1   C     ⁢     ∫       I   n     ⁢     ⅆ   t           =         L   m         R   1     ⁢   C       ·   ip           
 
 In this case, L m , the mutual inductance is well controlled, but can vary from part to part due to assembly variations. The capacitance C will vary from part to part and probably can be controlled to +/−5% accuracy. The capacitor  1214  will not have any appreciable temperature coefficient. R 1  is dominated by the metal resistance of the coil and will vary from part to part. It is equal to the value of the resistor  1210  and also has a large temperature coefficient. 
 
         [0040]     In order to obtain overall accuracy for the capacitance C which varies from part to part, a factory calibration using a one time programmable (OTP) memory  1220  can be used. In a preferred embodiment, a low cost 32 bit OTP memory may be utilized. The OTP memory  1220  provides a control variable to a programmable gain amplifier  1222 . The first gain stage  1223 , consisting of programmable amplifier  1222 , programmable resistance  1224  and the OTP memory  1220 , compensates for part to part variations of the circuit. The OTP memory  1220  is programmed at the factory based upon measurements made there. The programmable gain amplifier  1222  has its negative input connected to the output of the operational amplifier  1212 . A programmable resistance  1224  is connected between the output of the programmable amplifier  1222  and ground. The positive input of programmable amplifier  1222  is connected to the programmable resistance  1224 . The value of the programmable resistance  1224 , and thus the gain of the first gain stage  1223 , is controlled by the values provided from the OTP memory  1220 .  
         [0041]     A second gain stage  1226  compensates for differences in the resistance caused by temperature variations in the device. A temperature sensor  1228  and A-D converter  1230  are used to generate a digital temperature value to compensate for the coil resistance temperature coefficient. The temperature sensor  1228  detects the temperature and generates an analog representation of the temperature. The ADC 1230  converts the analog signal into a digital signal. The digital temperature value is provided via a control bus  1231  to control logic  1232 . In one embodiment the control logic  1231  may consist of a look-up table. The control table would include various control values associated with particular temperature values. Alternative embodiments may include a microprocessor programmed to control the output according to various temperature levels or other types of digital logic. The control logic  1232  provides a control value to the programmable gain amplifier  1234  and programmable resistance  1236 . The negative input of programmable amplifier  1234  is connected to the output of programmable amplifier  1222 . The programmable resistor  1236  is connected between the output of programmable amplifier  1234  and ground. The positive input of programmable amplifier  1234  is connected to the programmable resistance  1236 . The particular value of the programmable resistance  1236  , and thus the gain of the second gain stage  1226 , is controlled via the output from the control logic  1232 . The output of programmable amplifier  1234  provides the compensated V SENSE  signal. The code provided by the control logic  1232  is updated during the phase in which the operational amplifier  1212  is reset responsive to a reset signal applied to switch  1216  the reset signal is applied while the sensed current i p  is zero.  
         [0042]     The current sensor is designed to be used in, for example, a switched power supply. When the current i p  is equal to zero, a reset signal may be applied to switch  1216  to reset the capacitor  1214 , and the logic value applied to amplifier  1234  via control logic  1232  is updated responsive to the presently sensed temperature from temperature sensor  1228 . Referring now to  FIG. 13 , there is provided one example of how to apply the reset signal to a current sensor  1302  within a buck converter circuit. In this case, the buck converter circuit control signal φ 1  is applied to a transistor  1304  having its drain/source path connected between 12 volts and node  1306 . A second transistor  1308  has its drain/source path connected between node  1306  and node  1310 . Transistor  1308  is controlled by a second control signal φ 2 . The current sensor  1302  is connected between node  1310  and ground to detect current i p  and provide a control signal V SENSE . An inductor  1312  is connected between node  1306  and node  1314 . A capacitor  1316  is connected between node  1314  and ground. A load  1318  is also connected between node  1314  and ground. In one embodiment, the reset signal to switch  1216  of the current sensor  1302  may be configured to be the control signal φ 2 .  
         [0043]     As illustrated in  FIG. 14 , the current i p  is zero when signal φ 1  goes low and when signal φ 2  goes high at, for example, time t 1 . Integrator  1218  is reset during phase two when signal φ 2  goes high and the current sensor would accept signal φ 2  as an input to drive the reset signal to switch  1216 , since the current i p  is zero during this time. As can be seen each time the signal φ 2  goes high, the current i p  is zero enabling the reset signal to be applied to the integrator circuit  1218 .  
         [0044]     Referring now to  FIG. 15 , there is illustrated an alternative embodiment wherein the reset signal to the reset switch  1216  is generated responsive to a one shot circuit consisting of negative glitch detect circuit  1502  and one shot circuit  1504 . When the current i p  goes low as illustrated, for example, at t 1  in  FIG. 14 , the negative glitch detect circuit  1502  will detect the negative edge of current i p . In response to this detection, the negative glitch detect circuit  1502  generates a pulse to the one shot circuit  1504 . The one shot circuit  1504  then generates the reset signal to the reset switch  1216  responsive to the pulse from the negative glitch detect circuit  1502 . Other methods for detecting when the sensed current i p  goes to zero may also be utilized for generating the reset signal to reset switch  1216 . The examples illustrated in  FIGS. 13-15  are merely provided as examples of some embodiments thereof.  
         [0045]     Although the preferred embodiment has been described in detail, it should be understood that various changes, substitutions and alterations can be made therein without departing from the scope of the invention as defined by the appended claims.