Abstract:
A connection device for high frequency signals is disclosed. The connection device includes a printed circuit having an external face and a transmission line printed on the external face, and a coaxial connector surface mounted on the external face of the printed circuit. The transmission line is connected to the connector by means of a bump contact belonging to the transmission line. A central core of the connector is connected to the bump contact. The printed circuit has at least one internal ground plane disposed parallel to the external face and contributing to the matching of the transmission line. The internal ground plane defines a perforation therethrough that faces the bump contact.

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This application claims priority to foreign French patent application No. FR 09 04347, filed on Sep. 11, 2009, the disclosure of which is incorporated by reference in its entirety. 
     FIELD OF THE INVENTION 
     The invention relates to a connection device for high frequency signals comprising a printed circuit on one external face of which is printed a transmission line and a coaxial connector surface mounted on the printed circuit on the external face. The invention is of particular use for the transmission of radiofrequency signals in the X band, in particular for frequencies from 9 to 10 GHz. The invention can be applied to other frequency bands. 
     BACKGROUND OF THE INVENTION 
     In order to convey high frequency signals, a so-called micro strip technology implemented on printed circuit is, for example, used. Printed conductive tracts are produced on faces of the printed circuit. These faces may be external or internal and may be separated by one or more conductive planes or ground planes. The micro strip lines have particular dimensions such that, once associated with the conductive planes, they form impedance-matched lines. This matching makes it possible to provide a certain transparency of the line with regard to the signal conveyed. In other words, the aim is to minimize the electrical power losses of the signal along the line. 
     To convey the high frequency signals outside the printed circuit, coaxial connectors for connecting the micro strip lines to coaxial cables, whose impedances are also matched, are, for example, used. There are connectors intended for surface-mounting. This type of mounting does not require any metallized hole for a contact of the connector to be inserted into. More specifically, these connectors are connected to the printed circuit by bonding flat areas of the connector to external lands of the printed circuit. The connection may be made by means of a conductive glue or a solder paste interposed between the flat areas and the corresponding lands. The printed circuit on which the components are placed, such as the connectors, is immersed in a tank in which a hot fluid, generally in vapour phase, is used to solder the components. 
       FIG. 1  illustrates one embodiment of a printed circuit  10  intended to receive a connector  11  on one of its external faces  12 . This embodiment was implemented by the applicant without being disclosed. On this face, a transmission line  13 , produced in the form of a micro strip line, comprises a rectilinear part  14  terminated by a circular bump contact  15 , the diameter of which is greater than the width of the rectilinear part  14 . The width of the rectilinear part  14  is defined so that the transmission line has a determined impedance, for example 50 ohms. This impedance depends on the composition and the dimensions of the printed circuit  10 . For example, the printed circuit comprises an insulating substrate of relative permittivity ε r  equal to 3.38, the substrate separating the external face  12  from an internal ground plane situated facing the transmission line  13 . The internal ground plane is not visible in  FIG. 1 . To obtain an impedance of 50 ohms, a width for the rectilinear part  14  of 450 μm and a thickness of the substrate of 203 μm are, for example, selected. As for the diameter of the bump contact  15 , this is imposed by the diameter of the flat area of the core of the connector  11 . In  FIG. 1 , the core cannot be seen. It is concealed by the body  16  of the connector  11 . The body  16  is connected by flat areas  17  to a ground plane  18  produced on the external face  12  and surrounding the transmission line  13 . In the example shown, the diameter of the bump contact  15  is 1.7 mm. Metallized holes or vias  19  are used to connect the ground plane  18  to the internal ground plane. The ground plane  18  is positioned at a distance from the transmission line  13  so that its interaction with the transmission line  13  does not modify, or modifies only negligibly, the impedance of the transmission line  13  while providing shielding for the latter. The edges of the ground plane  18  are, for example, situated at a distance d of 500 μm from the transmission line  13 . The ground plane  18  notably allows for easier grounding for the surface-mounted components, in particular for the connector  11 . The ground plane  18  also provides strong electrical insulation between the various patterns printed on the external face  12 . 
       FIGS. 1   a  and  1   b  represent enlarged views of the transmission line  13  in its plane.  FIG. 1   a  represents a portion of the rectilinear part  14 . All along this part, the distance d is constant.  FIG. 1   b  represents the transmission line  13  at its bump contact  15 . The distance d, defined for the rectilinear part  14 , is maintained around the bump contact  15 . 
     Connecting the connector  11  with a multilayer structure, as illustrated in  FIG. 1 , produces spurious electrical effects, which in particular break the transparency of the connector with regard to the signal conveyed. This transparency is above all degraded at high frequency. These effects, notably linked to the intrinsic capacitive nature of the bump contact  15 , then result in mismatching and greater or lesser insertion losses on the signal. 
       FIG. 2  represents the matching level S 11  of the example represented in  FIG. 1  expressed in dB according to the frequency of the signal conveyed by the transmission line  13  and the connector  11 . The matching level illustrates the reflected electrical power. Still for this example,  FIG. 3  represents the insertion level S 21  or insertion losses expressed in dB according to the frequency of the signal. The insertion level illustrates the electrical power losses in transmission at the connection between the transmission line  13  and the connector  11 . 
     It is found, in this case, that this connection exhibits a matching level of −6 dB for a useful working frequency of 9.3 GHz, and insertion losses of 1.5 dB for this same frequency. Given these results, the connection is far from optimal. 
     To improve the transparency of this connection, the applicant has proposed correcting this mismatching. An exemplary embodiment of this correction is represented in  FIG. 4 . The transmission line  13  comprises, in the vicinity of the bump contact  15 , an inductive line section  20  and a capacitive line section  21  that make it possible to produce a low-pass type filtering element cooperating with the impedance added by the transition between the connector  11  and the bump contact  15 . The inductive line section  20  is formed by a micro strip line, the width of which is less than that of the rectilinear part  14 . The capacitive line section  21  is formed by a square-shaped bump contact, the side dimension of which is greater than the width of the rectilinear part  14 . The inductive line section  20  is positioned between the bump contact  15  and the capacitive line section  21 . In a first approach, the various electrical elements of the transition, associated with impedance values at a given frequency, form a π-form low-pass filter cell comprising an inductance, the inductive line section  20 , positioned between two capacitances, the capacitive line section  21  and the bump contact  15 . The dimension of the bump contact  15  is imposed by the connector  11  and the dimensions of the impedances  20  and  21  are matched to obtain the best transparency for the transition at the frequency concerned according to the characteristics of the printed circuit  10 , thickness and permittivity of the substrate. 
       FIG. 5  represents the matching level S 11  of the example represented in  FIG. 4  expressed in dB according to the frequency of the signal conveyed by the transmission line  13  and the connector  11 . Still for this example,  FIG. 6  represents the insertion losses expressed in dB according to the frequency of the signal. It is found, in this case, that this transition exhibits a matching level of approximately −18 dB for a useful working frequency of 9.3 GHz, and insertion losses of 0.27 dB for this same frequency. It is found that the correction made significantly improves the transparency of the transition. However, the results obtained exhibit a relatively narrow-band nature around the working frequency. It is therefore necessary to modify the design of the printed circuit for any change of working frequency, however small. Furthermore, the quality of the correction obtained in this way is highly sensitive to the production tolerances of the printed circuit and the positioning tolerances of the connector on the printed circuit. In other words, there is a risk of obtaining significant random effects on the quality of the correction between two printed circuits produced according to the same design. 
     SUMMARY OF THE INVENTION 
     The invention improves the transparency of a transition between a printed circuit and a coaxial connector for use at high frequency. The aim of the invention is to obtain the desired transparency for a significant band width about a given working frequency. 
     To this end, invention is, in one embodiment, a connection device for high frequency signals, comprising a printed circuit on one external face of which is printed a transmission line and a coaxial connector surface mounted on the printed circuit on the external face, the transmission line being connected to the connector by means of a bump contact belonging to the transmission line to which is attached a central core of the connector, the printed circuit comprising at least one internal ground plane parallel to the external face and contributing to the matching of the transmission line, wherein the ground plane is perforated by means of a resist facing the bump contact. 
     The surface-mounting of the core of the connector is done without metallized hole and therefore without connection to internal layers of the printed circuit. 
     By perforating the ground plane, the impedance added by the bump contact can be reduced. The transparency of the transition between the transmission line and the connector is thus enhanced. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The invention will be better understood, and other advantages will become apparent, from reading the detailed description of an embodiment given by way of example, the description being illustrated by the appended drawing in which: 
         FIG. 1  represents an exemplary embodiment of a printed circuit intended to receive a connector on one of its external faces; 
         FIG. 2  represents the matching level S 11  of the example represented in  FIG. 1 ; 
         FIG. 3  represents the insertion level S 21  or insertion losses of the example represented in  FIG. 1 ; 
         FIG. 4  represents an example of correction applied to the printed circuit of  FIG. 1 ; 
         FIG. 5  represents the matching level S 11  of the example represented in  FIG. 4 ; 
         FIG. 6  represents the insertion level S 21  or insertion losses of the example represented in  FIG. 4 ; 
         FIG. 7  represents, in cross section, an exemplary embodiment of a printed circuit according to the invention; 
         FIG. 8  represents, in perspective, the printed circuit of  FIG. 7 , complemented by an impedance correction; 
         FIG. 9  represents the matching level S 11  of the example represented in  FIG. 8 ; 
         FIG. 10  represents the insertion level S 21  or insertion losses of the example represented in  FIG. 8 ; 
         FIG. 11  represents, in perspective, a variant embodiment of a printed circuit according to the invention; 
         FIG. 12  represents the variant of  FIG. 11  seen perpendicularly to the printed circuit. 
     
    
    
     In the interests of clarity, the same elements are given the same identifiers in the various figures.  FIGS. 1 to 6  have already been described to introduce the invention. 
     DETAILED DESCRIPTION 
       FIG. 7  represents, in cross section, the printed circuit  10  containing, on its external face  12 , the transmission line  13  and its bump contact  15 . The internal ground plane here is given the identifier  25  and contributes to the matching of the transmission line  13 . According to the invention, the internal ground plane  25  comprises a resist  26  situated facing the bump contact  15 . In the absence of resist, the presence of the bump contact  15  facing the ground plane  25  creates a mismatching of the transmission line  13 . The resist makes it possible to reduce this mismatching. 
     Advantageously, the printed circuit  10  comprises a second ground plane  27 , parallel to the internal ground plane  25 . The internal ground plane  25  is situated between the transmission line  13  and the second ground plane  27  which may be positioned on a second external face  28  of the printed circuit  10 . The second ground plane  27  provides shielding for the matched assembly formed by the transmission line  13  and the internal ground plane  25 . The second ground plane  27  is continuous facing the bump contact  15 . In other words, the second ground plane  27  is not perforated facing the bump contact  15 . The second ground plane  27  therefore forms with the bump contact  15  a mainly capacitive impedance. This impedance, symbolized by a capacitor  29 , is less than that obtained in the absence of resist  26  between the ground plane  25  and the bump contact  15 . The presence of the ground plane  27  makes it possible to electromagnetically close the printed circuit  10  facing the bump contact  15  and therefore control the impedance of the connection of the connector  11  and of the transmission line  13  by means of the capacitor  29 . 
       FIG. 8  represents, in perspective, the transmission line  13  and the internal ground plane  25 . The bump contact  15  and the resist  26  are both circular and centred on one and the same axis zz′ perpendicular to the various planes of the printed circuit  10 . To avoid overloading the figure, the ground plane  18  is not represented. Only the vias  19  connecting the ground plane  18  to the internal ground plane  25  appear in  FIG. 8 . The vias  19  may also be connected to the ground plane  27  to impose one and the same ground reference locally at each via  19 . 
     Advantageously, the diameter of the resist  26  is greater than that of the bump contact  15  so as to limit the impedance generated between the bump contact  15  and the internal ground plane  25 . For example, for a bump contact  15  of 1.7 mm diameter, a resist  26  of 2.7 mm diameter is produced. Thus, in the impedance of the transition between the transmission line  13  and the core of the connector  11 , the interaction of the bump contact  15  and of the ground plane  27  is predominant relative to the interaction of the bump contact  15  and of the internal ground plane  25 . The diameter of the resist  26  should not, however, be too great to maintain its interaction with the transmission line in its rectilinear part  14 . 
     The impedance of the transition is reduced compared to that obtained in the embodiment of  FIG. 1 . However, the transparency is not perfect, and it is possible to correct it in the vicinity of the bump contact  15  by including in the transmission line an inductive line section  20  and a capacitive line section  21  in order to produce a low-pass type filtering cell cooperating with the impedance added by the connection between the central core of the connector  11  and the bump contact  15 . The identifiers  20  and  21  also appear in  FIG. 4 . In practice, the inductive  20  and capacitive  21  line sections fulfil the same function. It is, however, important to note that the impedance values added in  FIG. 8  by the two line sections  20  and  21  are less than those added in  FIG. 4 . This makes it possible to obtain a wider-band filtering for the embodiment of  FIG. 8 . This bandwidth is important to the reproducibility of the transparency of the transition. It has, in practice, been found that, with similar production tolerances for the printed circuit  10  and for the positioning of the connector  11  on the printed circuit  10 , the transparency of the transition defined in  FIG. 8  evolves within a range that is significantly less than for the transition defined in  FIG. 4 . 
     As an example,  FIG. 9  represents the matching level S 11  of the example represented in  FIG. 8  expressed in dB according to the frequency of the signal conveyed by the transmission line  13  and the connector  11 . Still for this example,  FIG. 10  represents the insertion losses expressed in dB according to the frequency of the signal. The numerical values proposed for  FIGS. 1 to 6  have been used again for the embodiment whose structure is represented in  FIGS. 7 and 8 . The printed circuit comprises two insulating substrates of relative permittivity ε r  equal to 3.38, the substrate separating the external face  12  from the internal ground plane  25  has a thickness of 203 μm and the substrate separating the internal ground plane  25  from the ground plane  27  has a thickness of 406 μm. 
     Two exemplary corrections are illustrated in  FIGS. 9 and 10  according to the length of the inductive line section  20 . In  FIG. 9 , a curve  9   a  represents the matching level for a length of the inductive line  20  of 1600 μm and a curve  9   b  represents the matching level for a length of 1200 μm. For a useful working frequency of 9.3 GHz, the matching level is approximately −26 dB in both correction examples. Similarly, for  FIG. 10 , a curve  10   a  represents the insertion losses for a length of 1600 μm and a curve  10   b  for a length of 1200 μm. Still for a useful working frequency of 9.3 GHz, the insertion losses are 0.15 dB for both correction examples. 
     The combined implementation of the resist  26  and of a low-pass type correction makes it possible to obtain an excellent level of performance for the transition between the connector  11  and the transmission line  13  over a very significant bandwidth. Obviously, the numerical examples proposed hereinabove should be adapted by tests according to the useful working centre frequency. 
     In the correction examples illustrated previously, the width of the inductive line section  20  is 200 μm. This width is smaller than the width of the rectilinear part  14  which is 450 μm in order to obtain an inductive effect. This width reduction also tends to oppose the passage of the signal in the transmission line  13 . This drawback can be limited by modifying the shape of the resist  26 . The bump contact  15  retains its circular shape centred on the axis zz&#39;, the shape being imposed by the choice of the connector  11 . A variant shape of resist  26  that makes it possible to limit this drawback is illustrated in  FIG. 11 . More specifically, the resist  26  comprises a circular part  30  centred on the axis zz′ and an elongate part  31  extending along a second axis  32  parallel to a main direction  33  of the rectilinear part  14  of the transmission line  13  in contact with the bump contact  15 . 
     This variant does not exclude implementing a correction using inductive  20  and capacitive  21  line sections. It has, however, been found that not locally having any ground plane under the inductive line section  20  greatly enhances the inductive nature sought for this element. Consequently, with a given inductive line width, the inductive line section  20  will, for example, be shorter than the case of  FIG. 8  and therefore of a more localized nature. The elongate part  31  can be extended only facing the inductive line section  20 . 
       FIG. 12  represents the variant of  FIG. 11  seen perpendicularly to the printed circuit  10 . Advantageously, a projection of the transmission line, on the internal ground plane  25 , maintains a distance d that is substantially constant relative to the edge of the resist  26 , the distance d being measured on the one hand between the bump contact  15  and the circular part  30  of the resist  26  and on the other hand between the rectilinear part  14  of the transmission line  13  and the edge of the resist  26  perpendicularly to the main direction  33  of the rectilinear part  14 . If the elongate part  31  extends only facing the inductive line section  20 , for this part  31 , the distance d is measured between the edge of the resist  26  and the inductive line section  20 . 
     It will be readily seen by one of ordinary skill in the art that embodiments according to the present invention fulfill many of the advantages set forth above. After reading the foregoing specification, one of ordinary skill will be able to affect various changes, substitutions of equivalents and various other aspects of the invention as broadly disclosed herein. It is therefore intended that the protection granted hereon be limited only by the definition contained in the appended claims and equivalents thereof.