Abstract:
An improved method of commutating an electronically commutated DC motor shuts off application of power between the end of one current pulse and the beginning of the subsequent current pulse. Based upon the instantaneous rotation speed, one calculates at what instant to shut off the power. During the power interruption, the disconnected winding is operated in short-circuit mode using two MOSFET transistors, and the decay of the current is monitored. When the current reaches a predetermined reduced value, the terminals of the winding are switched to a high-resistance state, until the subsequent current pulse is started. This has the advantage that less reactive power occurs during operation, and one need not install as bulky a storage capacitor as the capacitors used heretofore.

Description:
FIELD OF THE INVENTION 
   The invention concerns a method for commutating an electronically commutated DC motor, and a motor for carrying out such a method. 
   BACKGROUND 
   An electronically commutated motor “breathes” during operation; i.e. in addition to its normal energy consumption, it alternately receives energy from the power network or from a storage capacitor connected to the power network, and at intervals—during commutation—delivers energy to that storage capacitor. In electrical engineering, this phenomenon is also referred to as the phenomenon of reactive power. In order to adapt to this reactive power, the storage capacitor must be of considerable size (usually hundreds of μF) so it can temporarily store this energy. A storage capacitor of this kind has a limited service life and requires a great deal of space in the motor. 
   SUMMARY OF THE INVENTION 
   It is therefore an object of the invention to make available a novel method for commutating an electronically commutated motor, and a motor adapted to that method. 
   According to the invention, this object is achieved by executing, at the end of a current pulse, a switchoff routine designed to improve the conversion of electric energy, stored in the winding to be switched off, into mechanical energy driving the rotor. This is therefore a method for commutating an electronically commutated motor comprising an improved commutation procedure that is effective between the switching off of one current pulse and the switching on of a subsequent current pulse. In this, a switchoff time is calculated based on the instantaneous rotation speed, and energy delivery from a direct current source to the motor is interrupted when that time is reached. The winding to be switched off is then operated substantially in short-circuit by way of two bidirectionally conducting semiconductor switches, and the decaying current in the winding is monitored. When it has reached a reduced value, the winding terminals of the winding are temporarily switched to high resistance before the subsequent current pulse begins. The result is that during the commutation procedure, the energy stored electrically in the inductance of the motor is converted better into mechanical energy, so that less reactive power occurs and consequently a smaller storage capacitor is needed. 
   An advantageous motor for carrying out such a method employs a full bridge circuit comprising at least two field effect transistors (FET&#39;s) which serve to operate a motor winding in short circuit during a predetermined operating state of the motor, and an arrangement for monitoring the direction of the current flowing in said short circuit and to initiate opening of said short circuit when a reversal of said current direction is sensed. 

   
     BRIEF FIGURE DESCRIPTION 
     Further details and advantageous embodiments of the invention are evident from the exemplary embodiments, to be understood in no way as a limitation of the invention, that are described below and depicted in the drawings. In the drawings: 
       FIG. 1  is an overview circuit diagram of a preferred embodiment of a direct current motor according to the present invention; 
       FIG. 2  is a schematic diagram to explain the commutation sequence in a motor according to the existing art; 
       FIG. 3  is a schematic diagram similar to  FIG. 2 , to explain the commutation procedure in a motor according to the invention; 
       FIG. 4  is a diagram of the current as measured in the course of a commutation procedure; 
       FIG. 5  is a diagram of a current profile measured when a maximum current limiter is applied; 
       FIG. 6  is a state diagram to explain the invention; 
       FIG. 7  depicts the change in magnetic flux density in the rotor over a range of 360° el., and a motor current profile in which the current limiter is applied; 
       FIG. 8  is a depiction similar to  FIG. 7 , showing, in highly schematic fashion, the consequences of the application of current limitation because an adaptive controller, associated with the motor and correspondingly increasing the length BW of the current blocks, becomes effective; 
       FIG. 9  shows a specific exemplary embodiment comprising an Arizona Microchip microcontroller; this Figure shows a portion of the microcontroller&#39;s circuit, and that portion is not repeated in  FIG. 10  below; 
       FIG. 10  is a detailed circuit diagram of the hardware for generating signals Imin and Imax; 
       FIG. 11  is a detailed circuit diagram of the hardware for controlling H-bridge  137 ; 
       FIG. 12  is an overview diagram explaining the basic structure of the software that is used; 
       FIG. 13  is a flow chart indicating the basic execution of the program in motor  100 ; 
       FIG. 14  is a flow chart of the interrupt handler for detecting and servicing the various interrupts; 
       FIG. 15  is a diagram explaining the Figures that follow; 
       FIG. 16  is a flow chart of the Hall Interrupt routine that is executed upon occurrence of an edge of signal HALL; 
       FIG. 17  is a flow chart of the Imax interrupt routine that is executed at an edge of signal Imax; 
       FIG. 18  is a diagram explaining the response of the maximum current limiter when a motor is rotating rapidly; 
       FIG. 19  is a flow chart of the Imin interrupt routine that is executed upon occurrence of signal Imin; 
       FIG. 20  is a flow chart of the TIMEOUT interrupt routine; 
       FIG. 21  is a flow chart for incrementing pulse duty factor pwm; 
       FIG. 22  is a flow chart for decrementing pulse duty factor pwm; 
       FIG. 23  is a flow chart to explain the operations that occur during a commutation; 
       FIG. 24  is a flow chart to explain commutation at a normal rotation speed of motor  100 ; 
       FIG. 25  is a flow chart to explain details of the commutation procedure upon shutoff of a current in winding  102 ; 
       FIG. 26  is a diagram to explain operations during commutation; 
       FIG. 27  is a flow chart to explain the calculation of a time period t_HALL at low and high rotation speeds; 
       FIG. 28  is a diagram to explain the calculation of time period t_HALL at low and at high rotation speeds; 
       FIG. 29  shows a CALC_ACCEL routine for taking acceleration into account; 
       FIG. 30  shows an RGL routine for rotation speed regulation; 
       FIG. 31  shows a routine for adaptive modification of pulse duty factor pwm as a function of operating conditions of the motor; 
       FIG. 32  is a diagram to explain the mode of operation of  FIG. 31 ; and 
       FIG. 33  is a diagram with numerical values for a more detailed explanation of FIG.  26 . 
   

   DETAILED DESCRIPTION 
   In the description below, identical reference characters are used for identical or identically functioning parts, which are usually described only once. Since the subject is a difficult one, concrete numerical values—e.g. 3 A, 1.6 A, 200 μs, 1000 rpm, etc.—are often indicated in order to make the text more readable. It is understood as self-evident, however, that these concrete values are merely preferred examples which in no way limit the invention. 
     FIG. 1  provides an overview of a preferred embodiment of a motor according to the invention. 
   In this embodiment, the actual motor  100  has one winding phase  102  having two terminals  104 ,  106 , as well as a permanent-magnet rotor  108 . The exemplary embodiment below refers to a motor  100  having a four-pole rotor  108 , although any number of poles, and also other numbers of winding phases, are of course possible. The example of motor  100  was selected because of its simplicity, in order to facilitate comprehension of this very complex invention. 
   The exemplary embodiment shows a motor  100  in which a current i 1  flows from terminal  104  to terminal  106  in the region of a rotor rotation of 180° el., and a current i 2  flows from  106  to  104  in the region of the subsequent rotor rotation of 180° el. The duration (beginning and end) and amplitude of currents i 1  and i 2  are varied depending on the motor&#39;s needs; this is usually referred to as a so-called block control system, i.e. current i 1  can have a length e.g. of between 0° and 180° el., as can current i 2 . Also possible, without any additional effort, is a so-called “ignition advance,” as indicated in  FIG. 15  at VZ and explained in equations (3a) and (4a) below. 
   Since a motor of this kind requires only a single winding  102 , it is very simple. It is preferably used to drive fans. DE 2 346 380 and corresponding U.S. Pat. No. 3,873,897, Muller, give an example of the construction of such a motor, which is produced in a great many variants. 
   Motor  100  preferably has a galvanomagnetic rotor position sensor  110 , e.g. a Hall generator, controlled by rotor  108 , and that sensor is shown again on the left in FIG.  1 . Its output signals are amplified by an amplifier  112 , converted into square-wave HALL pulses, and then conveyed to a microcontroller μC  40 , where each edge of these HALL pulses triggers an interrupt (hereinafter referred to as a HALL interrupt) (cf. FIG.  16 ). Because of the magnetization of rotor  108 , a HALL interrupt of this kind is triggered each time rotor  108  has rotated through 180° el. The distance t_HALL between two HALL interrupts is large at low rotation speeds and small at high rotation speeds, and is therefore an indication of the rotation speed of rotor  108  which is used for rotation speed regulation (FIG.  30 ). The time span t_HALL corresponds to the time required by rotor  108  to rotate through 180° el.; cf. equations (6) and (7) below. 
   Terminal  104  of winding  102  is connected to drain D of a p-channel MOSFET  114  whose source S is connected to a positive line  116  that is connected via a protective diode  118  to a positive terminal  120  which usually is connected to a (schematically indicated) power supply unit  121  which supplies a DC voltage of, for example, 12, 24, 48 or 60 V depending on the type of motor  100 . The negative line (GND) of motor  100  is labeled  122 , and its terminal  124 . A capacitor  126  is located between positive line  116  and negative line  122 . 
   Motor  100  “breathes” as it operates, i.e. it alternately receives energy from power supply unit  121  and capacitor  126  and in the intervals during the commutation procedures—delivers energy, which must be temporarily stored by capacitor  126  so that the voltage between lines  116  and  122  does not become too high. Capacitor  126  therefore has a size that is approx. 500 μF in conventional motors with comparable performance data, and that can be significantly decreased in the context of the invention. It is not easy to accommodate large capacitors  126  in small motors. Because of the high temperature in a motor, the service life of such a capacitor is limited. One of the objects of the invention is therefore to keep capacitor  126  small and to place little electrical load on it. At an operating voltage of 12 V, for example, the size of this capacitor can be 60 to 100 μF when the motor is operating according to the invention. 
   Terminal  106  of winding  102  is connected to drain D of a p-channel MOSFET  130  whose source S is connected to line  116 . 
   Terminal  104  is also connected to drain D of an n-channel MOSFET  132  whose source S is connected via a measuring resistor  134  to negative line  122 . 
   Terminal  106  is likewise connected to drain D of an n-channel MOSFET  136  whose source is connected via a measuring resistor  138  to negative line  122 . 
   Free-wheeling diodes  114 ′,  130 ′,  132 ′,  136 ′ are connected in the usual manner antiparallel to MOSFETs  114 ,  130 ,  132 ,  136 . 
   Gate G of MOSFET  132  is connected to the output of an amplifier  140  to whose input  142  a signal LSL is conveyed from μC  40  when MOSFET  132  is to be switched on. (LSL is hereinafter also referred to as LSL_OUT, and similarly for signals LSR, HSL, and HSR.) 
   Gate G of MOSFET  136  is connected to the output of an amplifier  144  to whose input  146  a signal LSR is conveyed from μC  40  when transistor  136  is to be switched on. 
   Gate G of MOSFET  114  is connected to the output of an amplifier  148  whose input  150  is controlled by the output signal of a logic element  152 . Together with amplifier  148  this constitutes a NAND gate; i.e. when one of the input signals of logic element  152  is low, MOSFET  114  is blocked. In that case, logic element  152  has a low output signal. The resistance of driver amplifier  148  thus becomes high, pulling the potential at gate G of FET  114  upward so that the latter becomes nonconductive. 
   Gate G of MOSFET  130  is connected to the output of an amplifier  154  whose input  156  is controlled by the output signal of a logic element  160 . Together with amplifier  154  this constitutes a NAND gate; i.e. when one of the input signals of logic element  160  is low, MOSFET  130  is blocked. Because of the symmetry of the circuit, the mode of operation is the same as for FET  114 . 
   Both logic elements  152  and  160  have conveyed to them from μC  40  a PWM signal PWM which has e.g. a frequency of 20 kHz and whose pulse duty factor pwm can be set by means of μC  40  at between 0 and 100%. This signal PWM is continuously generated by μC  40  during operation, and determines the magnitude of the current conveyed to motor  100 . 
   Similarly, both logic elements  152  and  160 , as well as μC  40 , have a (low) signal Imax conveyed to them when the current in MOSFET  132  or in MOSFET  136  exceeds a defined limit value. This signal Imax results in immediate shutoff of both MOSFETs  114  and  130  by way of the motor&#39;s hardware. (Only one of these two MOSFETs  114 ,  130  can ever be active at any given point in time.) Signal Imax is therefore “low-active,” i.e. it shuts the current off when the signal becomes low. 
   Logic element  152  also has conveyed to it from μC  40  a commutation signal HSL for controlling transistor  114 . Similarly, logic element  160  has conveyed to it from μC  40  a commutation signal HSR for controlling transistor  130 . 
   The terms HSL, etc. are mnemonic and denote the following: 
   
     
       
             
             
             
             
           
         
             
                 
                 
             
           
           
             
                 
               HSL 
               High side left 
               Transistor 114 
             
             
                 
               HSR 
               High side right 
               Transistor 130 
             
             
                 
               LSL 
               Low side left 
               Transistor 134 
             
             
                 
               LSR 
               Low side right 
               Transistor 136 
             
             
                 
                 
             
           
        
       
     
   
   The four transistors  114 ,  130 ,  132 ,  136  constitute, together with winding  102 , a so-called H bridge (or full bridge)  137  having high-side (HS) transistors  114 ,  130  and low-side (LS) transistors  132 ,  136 . When transistors  114  and  136  are switched on, a current i 1  flows in winding  102  from left to right. When transistors  130  and  132  are switched on, a current i 2  flows in winding  102  from right to left. 
   Provided between inputs  142  and  150  is an interlock  166  which prevents transistors  114  and  132  from being conductive simultaneously. Similarly provided between inputs  146  and  156  is an interlock  168  which prevents both transistors  130  and  136  from being conductive simultaneously. These interlocks serve to protect H-bridge  137 . 
   The voltage at resistor  134  is conveyed through a signal filter  170  (to filter out interference pulses) to the positive input of a comparator  172  whose negative input is connected to a node  174  that is connected via a resistor  176  to negative line  122  and via a resistor  178  to a node  180  which is connected via a resistor  182  to a line  184  to which a regulated voltage of +5 V is applied. The voltage at resistor  176  thus constitutes a reference voltage Uref which determines the current at which the maximum current detector responds. 
   Node  180  is connected via a resistor  186  to the collector of an npn transistor  188  at which a (low) signal Imax is generated in the event of overcurrent, and which is therefore connected directly to logic elements  152  and  160  and to μC  40 , and also via a resistor  190  to line  184 . 
   The emitter of transistor  188  is connected to negative line  122 . Its base is connected via a resistor  191  to the cathodes of two diodes  192 ,  194  that are connected via a resistor  193  to negative line  122  (GND). The anode of diode  192  is connected to the output of comparator  172 . 
   The voltage at measuring resistor  138  is conveyed, via a signal filter  196 , to the positive input of a comparator  198  whose negative input is connected to node  174 . The output of comparator  198  is connected to the anode of diode  194 . 
   When the current through measuring resistor  134  becomes too high, the positive input of comparator  172  becomes more positive than the negative input, so that transistor  188  receives a base current through diode  192  and is switched on. When the current through resistor  138  becomes too high, the positive input of comparator  198  becomes more positive than its negative input, so that transistor  188  receives a base current through diode  194  and becomes conductive. 
   In both cases, resistor  186  is thereby switched in parallel with resistors  176 ,  178 , thereby increasing the current through resistor  182  and therefore the voltage drop at that resistor. As a result, reference voltage Uref automatically drops as soon as transistor  188  switches on, and this causes a switching hysteresis, i.e. comparator  172  switches on e.g. at an overcurrent of 3 A, and shuts off again only at approximately 1.6 A, and likewise for comparator  198 . This means that high-side transistors  114 ,  130  are forced to switch off e.g. at 3 A and can be (but do not need to be!) switched back on only when the current in resistor  134  or  138  has dropped to 1.6 A. This prevents overloading of high-side transistors  114 ,  130 , i.e. in the event of an overcurrent the presently conductive transistor is completely shut off as soon as the low signal Imax is generated at the collector of transistor  188 , and it cannot be switched back on until signal Imax is no longer being generated and the other criteria for switching it on are present, as will be explained in more detail below. 
   A comparator  202 , whose negative input is connected to the positive input of comparator  172  and whose positive input is connected to the positive input of comparator  198 , serves to recognize the zero transition for the instance in which both high-side transistors  114 ,  130  are blocked and both low-side transistors  132 ,  136  are conductive. 
   When the two low-side transistors  132 ,  136  are made conductive after shutoff of a previously conductive high-side transistor  114  or  130 , the current generated by the electrical energy stored in winding  102  causes a voltage drop at both resistors  134 ,  138 ; and when the current through winding  102  transitions from motor mode into generator mode, as is the case in  FIG. 3  at point  222 , this current changes direction and passes through zero. 
   For example, when current is flowing in motor mode from terminal  106  through resistors  138 ,  134  to terminal  104 , the positive input of comparator  202  is more positive than its negative input. After the zero transition, current flows from terminal  104  through resistors  134 ,  138  to terminal  106 , and the negative input of comparator  202  now becomes more positive than the positive input, so that at the current&#39;s zero transition, signal Imin at the output of comparator  202  abruptly changes, i.e. either from low to high or from high to low. At the zero transition an abrupt signal change (switching edge) thus occurs at the output of comparator  202 , and this brings about an interrupt in μC  40  that causes all four transistors  114 ,  130 ,  132 ,  136  to be blocked. This interrupt is referred to as an “Imin interrupt” and will be explained in more detail in  FIG. 19  below. 
   To explain the general mode of operation of  FIG. 1 , reference will be made to  FIGS. 2 and 3 , which explain the operating principle in highly schematic fashion.  FIG. 2  shows the current profile in the stator for a motor according to the existing art, and  FIG. 3  shows the analogous profile for a motor according to the invention.  FIGS. 2 and 3  show, over a rotation angle of 360° el., the following values: 
   a) Magnetic flux density B at rotor  108   
   Magnetic flux density is measured in tesla (T). Its profile in this example is approximately trapezoidal, and the term “trapezoidal magnetization” is therefore used. This is a preferred profile of B in the context of the present invention, but not the only conceivable one. 
   The changes in magnetic flux density B induce a voltage in stator winding  102  when rotor  108  rotates. The shape of this voltage corresponds to the shape of B, i.e. is also trapezoidal in this case. The amplitude of this voltage increases with increasing rotation speed. This voltage is referred to as the “induced voltage” or “counter-EMF.” 
   b)  FIG. 2  shows the stator current profile in a conventional motor Current i 1  through winding  102  usually begins at a time after 0° el. and rises rapidly at first (at  210 ) because of the low value of B, i.e. the low counter-EMF, in this region. The result of this rise is that some of the energy conveyed by current i 1  is transformed, with a time delay, into kinetic energy of rotor  108 . Current i 1  then decreases again slightly at  211 , because of the higher counter-EMF, to a minimum  212 . In  FIG. 2 , i.e. in a conventional motor, current i 1  rises from  212  to a maximum  216  where current i 1  is shut off, and then drops to zero along a curve  218 . In this example (FIG.  2 ), zero transition  217  is reached somewhat before 180° el., but can also occur after 180° el. depending on the angular position of Hall generator  110 . 
   Because of the symmetry of the arrangement, the events for current i 2  that flows from terminal  106  to terminal  104  are analogous and are therefore not described again. In  FIG. 2 , current i 2  begins at 180° el. 
   Time span P between point  217  and the onset of current i 2  is referred to as the switching off-time or switching gap P. This is necessary, among other reasons, in order to prevent a short circuit in H bridge  137 . (For example, if transistors  114  and  132  in  FIG. 1  were conductive simultaneously, a short-circuit current would occur through them from positive line  116  to negative line  122 .) 
   In an ECM with conventional commutation, in the angular range approximately from 0° el. to maximum  216  the winding current i 1  is converted with a time delay into kinetic energy of rotor  108 . 
   When current i 1  is abruptly switched off at point  216 , a high induced voltage occurs at winding  102  and attempts to make that current i 1  continue to flow, so that between points  216  and  217  current i 1  flows through freewheeling diodes  132 ′ and  130 ′ to capacitor  126  and charges it. Energy E stored in winding  102  is transferred almost entirely into capacitor  126 , meaning that the latter must be very large so that the voltage between lines  116  and  122  does not rise excessively. Energy E depends on the square of current I at time  216 , and on inductance L of winding  102 :
 
 E=I   2   *L/ 2  (1)
 
where
 
   E=magnetic energy stored in winding  102 ; 
   I=instantaneous current in winding  102 ; 
   L=inductance of winding  102 . 
   Since I is very high at shutoff, energy E that is stored inductively in winding  102  is also very high. 
   After the shutoff of winding  102 , this energy is transferred into capacitor  126 . This is therefore a reactive power component that shuttles back and forth between capacitor  126  and winding  102 ; and because this reactive power is high, capacitor  126  must also be large. The large currents that flow as a result of this reactive power also cause unnecessary losses that reduce the motor&#39;s efficiency. 
   It is the intent of the invention to reduce this reactive power, i.e. to have as little energy as possible flow out of winding  102  into capacitor  126  at shutoff, but instead to drive rotor  108  using that energy. 
   The Commutation Procedure According to the Invention ( FIG. 3 ) 
   A commutation procedure that differs greatly from the conventional one is therefore used (as shown in FIG.  3 ). In  FIG. 3 , current i 1  once again rises sharply at  210  after switching on, and decreases at  211 . To that extent the profile is similar to that in FIG.  2 . It is different in the following ways, however: 
   a) Energy delivery from lines  116 ,  122  to winding  102  is shut off at a point  214  calculated by μC  40 , usually at a point where motor current i 1  has not yet reached its maximum  216  (FIG.  2 ). The calculation of shutoff time  214  is described in FIG.  30 . Shutoff is effected by shutting off, at point  214 , the instantaneously conducting high-side transistor (either  114  or  130 ).  FIG. 25  below describes, by way of example, how this can be done. 
   b) Subsequent to point in time  214 , usually after a short off-time, both low-side transistors  132  and  136  are then made conductive (cf.  FIG. 25 , S 840 ) so that current i 1  can continue to flow through these two transistors; in FET  136  it flows from drain D to source S, which is possible in a FET. This results in a low-resistance connection between terminals  104  and  106  of winding  102 , and in this connection current i 1  decays along a curve  220 , continuing to drive rotor  108  (i.e. to generate motor-mode energy). 
   c) At a point  222 , current i 1  transitions through zero and would thereafter continue to flow as generator-mode current  224  if transistors  132  and  136  were to continue conducting. This current  224  is indicated as a dotted line. Since it would have a braking effect, it is undesirable. 
   To prevent this, OP amplifier  202  ( FIG. 1 ) generates signal Imin in the vicinity of point  222 . This signal generates an Imin interrupt in μC  40 , so that the latter immediately makes all four transistors  114 ,  130 ,  132 ,  136  of H-bridge  137  nonconductive. In the example shown in  FIG. 3 , this occurs shortly after point  222 . 
   Since current i 1 =0 at time  222 , no more energy is stored in winding  102  when all the MOSFETs are shut off. As a result, after the shutoff of winding  102  no energy can be fed back from it into capacitor  126 . 
   All that is still present at winding  102  at this point in time is the voltage induced by rotor magnet  108 ; but this is low at time  222  (usually amounting to only a few volts) and is therefore unproblematic. 
   After a short switching off-time P 1 , current i 2  is then switched on. The switch-on time is calculated by μC  40  (cf. FIG.  30 ). 
   At motor start-up it would take too long for i 1  to reach a value of zero in segment  220 , so here the current is shut off by a special function (called the TIMEOUT function) after a predefined time, e.g. after 500 to 800 μs, even if i 1  (or i 2 ) has not yet reached a value of zero. The time T 3  elapsed after reaching point  214  at which high-side transistors  114 ,  130  are shut off is therefore monitored here, as is current Imin. All the transistors of H-bridge  137  are shut off no later than the point at which T 3  elapses, or alternatively upon generation of the Imin interrupt, if that occurs earlier than the end of T 3 . T 3  is typically in the range from 500 to 800 μs. 
     FIG. 4  shows the current through winding  102  that is actually measured during operation and, for comparison, current I in supply lead  116  (FIG.  1 ). The current through winding  102  changes direction as rotor  108  rotates, while current I flows in only one direction. For better comparison, current I is plotted downward from a zero line  98 . 
   Current i 2  receives its shutoff command at a time t 10  in this case, so that high-side transistor  130  is blocked and, after a short delay, both low-side transistors  132 ,  135  are switched on, causing current i 2  to decay along a curve  220 A. 
   Current i 2  passes through zero at a time t 11 , and at a time t 12  the Imin interrupt (already described) becomes effective, causing all four transistors  114 ,  130 ,  134 ,  136  to be blocked so that no current flows in winding  102  from a time shortly after t 12  until a time t 13 . 
   At time t 13  which is calculated in μC  40  (cf. FIG.  30 ), current i 1  is switched on by making transistors  114  and  136  conductive, so that current i 1  rises as depicted. At a time t 14  that is calculated in μC  40 , i 1  is shut off by blocking high-side transistor  114  and making both low-side transistors  132 ,  136  conductive, so that current i 1  decreases along a curve  220 B and reaches a value of zero at time t 15 . Shortly thereafter, the Imin interrupt takes effect and blocks all four transistors  114 ,  130 ,  132 ,  136  until a time t 16  at which transistors  130  and  132  are switched on so that current i 2  can flow. 
     FIG. 4  shows that to the left of t 10 , current I in supply lead  116  is identical to current i 2  in winding  102 . 
   At time t 10 , current I can no longer flow out of positive line  116  because high-side transistor  130  is open and the two low-side transistors  132 ,  136  are conductive, so that current i 2  continues to flow only through these two transistors. From t 10  to t 13 , the value of current I therefore remains practically at zero. 
   From t 13  until t 14 , the profile of I is the mirror image of i 1 , i.e. the two currents are identical in magnitude. From t 14  to t 16 , I has a value of zero, and after t 16  I once again has practically the same value as i 2 , although some additional energy may possibly be conveyed out of capacitor  126  shortly after t 16 . 
   The invention therefore largely prevents energy from shuttling back and forth between winding  102  and capacitor  126 , so that the dimensions of capacitor  126  can be correspondingly smaller. 
     FIG. 5  shows, on an oscillogram, a typical profile of the currents that occur when the current limiter takes effect. This limits currents i 1  and i 2 , in this exemplary embodiment, to a value Imax=3 A. 
   Current i 1  begins at t 20 . The commutation control system in μC  40  causes current i 1  to be interrupted at a time t 21  by the opening of transistor  114 ; and from t 21  until a time t 22 , winding  102  is short-circuited because both transistors  132 ,  136  are conducting. 
   At t 23  transistors  130 ,  132  are switched on so that a current i 2  flows. 
   This current rises rapidly to the negative current limit value −Imax. There, at time t 24 , high-side transistor  130  is blocked by signal Imax, so that current i 2  drops until a time t 25 , both transistors  132 ,  136  being made conductive. At t 25 , transistor  188  shuts off signal Imax again because i 2  has dropped to 1.6 A, so that i 2  once again rises because transistor  130  is once again conductive. 
   At a time t 26  the commutation control system opens transistor  130 , and both low-side transistors  132 ,  136  are switched on so that i 2  reaches a value of zero at t 27 . At t 28 , i 1  is switched on again by making transistors  114  and  136  conductive. 
   Each time signal Imax becomes low, pulse duty factor pwm of signal PWM is reduced slightly (cf. S 508  in  FIG. 17 ) so that after a few revolutions the values +Imax and −Imax are no longer reached and the “smooth” current profile shown in  FIG. 3  is once again obtained. While the maximum current is being lowered below Imax (3 A), value BW is increased by the controller (FIG.  30 ), if possible, and pulse duty factor pwm is also, if applicable, slowly raised until the motor is once again running normally, i.e. at the desired rotation speed. The operation may possibly also repeat, i.e. signal Imax may occur again, if pwm is raised excessively. 
   In  FIG. 5 , the points at which the commutation control system interrupts the relevant current are labeled t 29  through t 37 . The values at which current limiting is applied are labeled +Imax and −Imax, and the current values resulting from switching hysteresis are labeled +ImaxHY and −ImaxHY. In the exemplary embodiment, Imax=3 A and ImaxHY=1.6 A. 
     FIG. 6  once again shows the operations just described, in graphic fashion with reference to a state diagram. At  230 , motor  100  is in region  210 ,  211  of  FIG. 3 , and the system monitors whether point  214 , at which energy delivery from lines  116 ,  122  to motor  100  needs to be terminated, has been reached, 
   If it is determined at  230  that the end of current flow has not yet been reached, current flow is then continued in state  234 , and monitoring then continues at  230  to determine whether time  214  has been reached. If so, motor  100  then enters state  236  HS OFF, in which both high-side transistors  114 ,  130  are shut off, interrupting energy delivery to motor  100 . 
   The program then enters a short DELAY  238  and then, in state LS ON  240 , switches on both low-side transistors  132 ,  136  so that winding  102  is essentially operated in short circuit and the current decays along curve  220  (FIG.  3 ). This is monitored in the next state  242  (“Wait until current has dropped to zero”), while the current in winding  102  continues to drive rotor  108 . 
   When the current reaches a value of zero, comparator  202  generates a signal Imin and causes an Imin interrupt  244 . 
   Simultaneously, at  246  the TIMEOUT function monitors whether the predefined time T 3  ( FIG. 3 ) has elapsed. 
   The earlier of the two events (TIMEOUT  246  or Imin interrupt  244 ) causes the transition to state  248 , i.e. complete shutoff of all four transistors of H-bridge  137  (LS OFF &amp; HS OFF). In this state, the kinetic energy of rotor  108  cannot be transported in generator mode into capacitor  126 , since the instantaneous value of the voltage produced in generator mode by rotor  108  is lower than the voltage between lines  116  and  122 . 
   The “breathing” of motor  100  described initially is therefore largely suppressed here by skillful energy management, i.e. during normal operation of motor  100 , very little reactive power flows back and forth between winding  102  and capacitor  126 . Because of the duration of the requisite calculation steps, however, Imin interrupt  244  cannot be generated exactly at the zero transition time  222  ( FIG. 3 ) but instead only slightly thereafter, and therefore a capacitor  126  is still necessary for temporary storage of energy from the motor, although it can be smaller than before. This capacitor is also needed in order to absorb energy when the motor is shut off, and to prevent an excessive rise in the voltage between lines  116  and  122 . 
   Maximum Current Detection Function 
   Maximum current detection by means of comparators  172  and  198  has already been described in  FIGS. 1 ,  3 ,  4 , and  5 . This function generates signal Imax, which acts via logic elements  152 ,  160  ( FIG. 1 ) directly on high-side transistors  114 ,  130  and, in the event of overcurrent, immediately shuts off transistor  114  or  130  that is conductive at that moment. In addition, signal Imax is also conveyed to μC  40  and generates an Imax interrupt there. The result of this, inter alia, is to initiate program steps that, in the context of subsequent current pulses, lower the current through winding  102  sufficiently that overcurrent no longer recurs. 
   Specifically, if the current through measuring resistors  134 ,  138  exceeds a value set at resistor  176  (referred to in  FIG. 1  as Uref), an Imax interrupt is therefore generated in μC  40  and high-side transistors  114 ,  130  are shut off directly by hardware. After a short delay has elapsed, both low-side transistors  132 ,  136  are switched on so that terminals  104 ,  106  of winding  102  are short-circuited through the two FETs  132 ,  136 . The next program steps depend essentially on the type of motor and its rotation speed, i.e. several variants are possible. 
   In one variant, when the current in winding  102  reaches a value of zero, Imin interrupt  244  is generated in the manner already described. For safety&#39;s sake, the time since LS ON  240  ( FIG. 6 ) is additionally measured by means of TIMEOUT function  246  (already described). 
   If the TIMEOUT time expires before Imin interrupt  244  is generated, this causes an OFF command for both low-side transistors  132 ,  136 . If the Imin interrupt occurs first, it causes the LS OFF signal. After a delay, current flow through winding  102  is then continued; i.e. if, in the instantaneous rotational position of rotor  108 , the current in winding  102  should be flowing from  104  to  106 , transistors  114 ,  136  are switched back on and transistors  130 ,  132  remain shut off. For a current in the opposite direction (from  106  to  104 ), the converse applies accordingly. 
     FIG. 7  schematically shows current pulses i 1 , i 2  whose amplitude A 1  reaches that of current Imax (3 A) at points  250 ,  251 , so that at these points the current limiter takes effect and the current drops until a point  252  or  253  is reached. There the current is switched back on because the (low-active) signal Imax is no longer being generated, and the current rises again until points  255  or  257 , where the shutoff command is issued by μC  40 . At both points  250  and  251 , pulse duty factor pwm is reduced by program step S 508  of  FIG. 17  in order to reduce amplitude A 1 . 
   As shown in  FIG. 8 , the result of this reduction of pulse duty factor pwm is that, after a time delay, amplitude A 2  of the current in motor  100  is reduced to a value that is less than 3 A, as symbolized by white arrows  254 ,  256  of FIG.  8 . In  FIG. 7 , the block length of a pulse—namely the time from the switch-on command to the switch-off command—has the value BW 1 . 
   As compensation for the reduction in amplitude from A 1  to A 2 , in  FIG. 8  the block length BW for controlling pulses i 1 , i 2  is extended to a value BW 2 , as symbolically indicated by black arrows  258 , so that there is no change in the energy delivered to motor  100 , i.e. area F 1  under curve i 1  of  FIG. 7  corresponds substantially to area F 2  under curve i 1  of FIG.  8 . To explain this in illustrative terms, in  FIG. 8  a force  254 ,  256  slightly widens the width of pulses i 1 , i 2  so that amplitude A 1  is no longer reached, the lower amplitude A 2  of currents i 1 , i 2  being compensated for in  FIG. 8  by increasing their block length BW 2 . 
   This is important because losses resulting from the processes described in  FIG. 5  increase when the maximum current is exceeded, and there is a risk of overloading the MOSFETs. Motor  100  also runs more quietly when it is operated at a current below its preset maximum current. Of course block length BW attained by pulses i 1  and i 2  must always be slightly less than 180° el., since otherwise a bridge short circuit might occur. 
   If block length BW of pulses i 1 , i 2  becomes too long, it is shortened by the motor&#39;s software; and as compensation in such a case, the amplitude is increased, i.e. the motor then tends to go from the state shown in  FIG. 8  to the state shown in FIG.  7 . In such a case the direction of arrows  254 ,  256 ,  258  is reversed. 
   At startup, the startup current may optionally be limited by the current limiter, but it is also possible to start up without overcurrent by slowly increasing pulse duty factor pwm of signal PWM ( FIG. 1 ) in ramped fashion. 
   In order to implement the invention, the motor&#39;s software calculates: 
   a) the pulse duty factor pwm that signal PWM should have at each moment; 
   b) the time at which a current pulse must be switched on; and 
   c) the time at which a current pulse must be shut off. 
   This is explained below. 
   In the exemplary embodiment, block length BW is calculated by a rotation speed controller that is described below in FIG.  30 . BW is thus predefined for calculation purposes, and is independent of pulse duty factor pwm of signal PWM. (The pulse duty factor can, of course, also be completely or partially taken into consideration in calculating BW, but omitting such consideration makes the program shorter, which is important in a motor.) 
     FIG. 9  shows a portion of the circuitry of microcontroller (μC)  40  used in the exemplary embodiment, in this case a PIC16C72A of Arizona Microchip. This operates here at a clock frequency of 4 MHz. It has 28 inputs 1 through 28, designated as follows:
     1 MCLR/ (reset input)   2 through  7  RA 0 -RA 5     8 VSS (ground terminal)   9 CLKIN   10 CLKOUT   11 through  18  RC 0 -RC 7     19 VSS 1  (ground terminal)   20 VDD (+5 V)   21 through  28  RB 0 -RB 7     
   Terminals RA 1  through RA 5 , RC 3 , RC 4 , and RB 1  through RB 5  are each connected via a resistor R (10 kilohm) to ground GND, since these terminals are not used. These resistors are not depicted in  FIG. 10  to enhance the clarity of that depiction. 
   Terminals CLKIN and CLKOUT are connected to a quartz oscillator  42 . Terminals VSS and VSS 1  are connected to ground, and terminal VDD to a positive line at +5 V (regulated). A filter capacitor  44  (e.g. 100 nF) is present between terminals VDD and VSS. 
   Reset input MCLR/ is connected via a resistor  46  to a node  48  that is connected via a resistor  50  to +5 V and via a capacitor  52  to GND. Capacitor  52  is discharged upon startup, so that input MCLR/ then has a potential of 0 V, triggering a reset operation at startup. Capacitor  52  then charges through resistor  50  to 5 V. 
   RA 0  is the input of an A/D converter internal to μC  40 . A voltage between 0 and 4.5 V (Vcc) can be conveyed to this input, and is converted into a digital signal. The signal at RA 0  corresponds to the desired rotation speed. It is conveyed to an input  261  as PWM signal  262 , whose pulse duty factor pwm contains the rotation speed information. 
   A comparator  264  serves to process PWM signal  262  and standardize it to a regulated amplitude a. Its positive input is connected to a node  266  that is connected via a resistor  268  to a regulated +5 V voltage which is also supplied to μC  40 , and via a resistor  270  to GND. Resistors  268 ,  270  are selected so that a potential of +2.3 V is present at node  266 . 
   The negative input of amplifier  264  is connected to a node  272  that is connected via a resistor  274  to input  261  and via a resistor  276  to GND. Resistors  274 ,  276  can be of identical size. 
   Output  278  of amplifier  264  is connected via a pull-up resistor  280  to +5 V and via a resistor  282  to RA 0 . A capacitor  284  is present between RA 0  and GND. Components  282  and  284  together constitute a lowpass filter. 
   Signal  262  is inverted by amplifier  264  to yield signal  286  at output  278 , which has a constant amplitude a, and that signal  286  is smoothed by lowpass filter  282 ,  284  to produce a DC voltage which is conveyed to input RA 0  and converted there, at each request, into a digital value. Since signal  286 , unlike signal  262 , has a defined amplitude a, its pulse duty factor is converted into a defined DC voltage and into a defined digital value. 
   Alternatively, the signal at input RA 0  can be generated in any other manner, e.g. by means of a potentiometer. In this processor, the maximum amplitude at RA 0  corresponds to 5 V. This corresponds to the internal A/D reference. 
   μC  40  has a ring counter TIMER 1  as well as a RAM and a ROM. An external RAM, EEPROM, or the like can additionally be provided, as is self-evident to one skilled in the art. 
     FIGS. 10 and 11  show a detailed exemplary embodiment of the circuit in FIG.  1 .  FIG. 10  shows the hardware for detecting Imax and Imin, as well as Hall generator  110 .  FIG. 11  shows μC  40  and H-bridge  137  that it controls. Parts identical to, or having the same function as, parts in the previous Figures are labeled with the same reference characters as therein, and usually are not described again. 
   The transitions from  FIG. 10  to  FIG. 11  are labeled  290 ,  292  (for H-bridge  137 ),  294  for signal HALL,  296  for signal Imin, and  298  for signal Imax. These are also shown in FIG.  1 . 
     FIG. 10  shows Hall generator  110 , whose output signal is amplified by means of a comparator  300  whose output  294  is connected via a pull-up resistor  302  to positive line  43  (+5 V, regulated). The square-wave HALL signals are conveyed to input RB 0  of μC  40 , where each edge of this signal causes a Hall interrupt (cf. FIG.  16 ). Hall generator  110  is supplied with current from line  43  through a resistor  304 . 
   The positive input of comparator  172  is connected via a resistor  305  to its output  307 , via a resistor  306  to node  290  and additionally to the negative input of comparator  202 , and via a capacitor  308  to GND. Resistor  306  and capacitor  308  together constitute lowpass filter  170  of FIG.  1 . Output  307  is connected via a resistor  309  to positive line  43 . 
   Similarly, the positive input of comparator  198  is connected via a resistor  309  to its output  311 , via a resistor  310  to node  292  and to the positive input of comparator  202 , and via a capacitor  312  to GND. Resistor  310  and capacitor  312  together constitute lowpass filter  196  of FIG.  1 . Output  311  is connected via a resistor  314  to positive line  43 . 
   The negative inputs of comparators  172 ,  198  are connected to node  174 , at which reference potential Uref is present at resistor  176 . 
   The positive input of comparator  202  is connected via a resistor  316  to its output  318 , which is connected via a resistor  320  to positive line  43 . 
   Signal Imin is obtained at output  318  of comparator  202 . It is conveyed through a resistor  297  to port RB 7  of μC  40 . Output  318  changes its potential at the zero transition of the motor current, as already described, and the switching edge at the transition causes an Imin interrupt in μC  40  (cf.  FIG. 19  below). 
   When, as a result of a stator current of e.g. 3 A, the voltage drop at resistor  134  becomes greater than voltage Uref at resistor  176 , the output of comparator  172  becomes high-resistance and acquires a high potential. As a result, a base current flows through resistor  309  and diode  192  to transistor  188  and makes the latter conductive, so that signal Imax at node  298  becomes low and thereby reduces the potential at nodes  180  and  174 . This implements the switching hysteresis already described, i.e. voltage Uref becomes correspondingly lower so that signal Imax becomes high again only when the current in resistor  134  has dropped to, for example, 1.6 A. The cathodes of diodes  192 ,  194  are connected via a common resistor  193  to GND. 
   Because of the symmetry of the arrangement, the same applies when the stator current through resistor  138  exceeds a value of 3 A. In this case as well, transistor  188  becomes conductive, implements the aforementioned switching hysteresis, and generates a low signal Imax at terminal  298  which does not become high again until that current has dropped to, for example, 1.6 A. 
   As shown in  FIG. 11 , signal Imax is conveyed directly to logic elements  152  and  160 , and by way of them blocks high-side MOSFETs  114 ,  130 . It is also conveyed, via a resistor  324 , to input RB 6  of μC  40 . As a result, both signals HSL and HSR are switched to low, so that one of the high-side transistors  114 ,  130  can switch back on only when
     a) signal Imax has once again become high; and   b) the associated signal HSL or HSR has also once again become high.   

   The results of this are as follows:
         Upon generation of signal Imax, e.g. at a current of 3 A, high-side transistors  114 ,  130  are blocked directly by the hardware and, shortly thereafter, additionally by μC  40 .   After signal Imax has ended, μC  40  can retain control over highside transistors  114 ,  130  and, for example, continue to block them if time BW ( FIGS. 7 and 8 ) has elapsed.       

     FIG. 11  shows that logic element  152  has a node  326  that is connected via a resistor  328  to port RC 0  of μC  40  and receives from there signal HSL for commutation. Also connected to node  326  are the anodes of three diodes  330 ,  331 ,  332 . The cathode of diode  330  is connected to port RC 2 , at which a PWM signal PWM (20 kHZ), whose pulse duty factor pwm is modifiable by means of software commands, is continuously generated. The cathode of diode  331  is connected to node  298 , to which signal Imax is conveyed. The cathode of diode  332  is connected to the base of npn transistor  148  and via a resistor  334  to GND. The emitter of transistor  148  is connected to GND, and its collector is connected via a resistor  336  to gate G of MOSFET  114 . The latter is connected via resistor  338  and a capacitor  340  parallel thereto to line  116 , i.e. to the operating voltage of motor  100 , which is also referred to as the DC link voltage. 
   As long as diodes  330 ,  331  are not conductive, and a high signal HSL is being conveyed from port RC 0 , node  326  has a high potential and diode  332  is conductive and conveys a base current to transistor  148  so that the latter is conductive and a current flows through resistors  338 ,  336 , thus generating at gate G of transistor  114  a signal that is a few volts more negative than the signal at its source S, so that transistor  114  is completely switched on. Capacitor  340  causes a slight delay in the switching operations and prevents oscillations. 
   The cathode of interlock diode  166  also receives GND potential, so that gate G of MOSFET  132  cannot have any positive potential conveyed to it in order to switch it on; in other words, transistors  114 ,  132  are interlocked with respect to one another. 
   When the potential of node  326  becomes low, for example because one of diodes  330 ,  331  becomes conductive or is receiving a low signal HSL from port RC 0 , diode  332  is blocked so that transistor  148  no longer receives base current and is also blocked. As a result, gate G of MOSFET  114  receives, through resistor  338 , the potential of positive line  116 , so that MOSFET  114  is blocked. The cathode of interlock diode  166  thereby receives a high potential, so that low-side MOSFET  132  can now be switched on. 
   Signal LSL is conveyed from port RC 6  via a resistor  342  to the base of npn transistor  140 . As long as this signal is high, or the cathode of interlock diode  166  is at a low potential, a low potential is present at the collector of transistor  140  and is conveyed via a resistor  346  to the gate of MOSFET  132  and blocks it. This gate is connected via a capacitor  348  to GND in order to delay the switching operations slightly. 
   When signal LSL at port RC 6  is low, transistor  140  is blocked. If the potential at the cathode of diode  166  is high, a high potential is now obtained via resistor  344  at the collector of transistor  140 , and this, via resistor  346 , makes MOSFET  132  conductive. The gate of MOSFET  132  is connected via a resistor  350  and a diode  352  to the anode of diode  166 , and when the latter&#39;s cathode is at GND, a positive potential at the gate of MOSFET  132  is immediately discharged through resistor  350 , diode  352 , and diode  166  to GND, so that MOSFET  132  becomes blocked. Since resistor  350  is preferably smaller than resistor  346 , the ratio between charging time constant and discharging time constant can be varied. These constants are also a function of the gate capacitance and other capacitances in the circuit. 
   The right half of the circuit shown in  FIG. 11  is entirely symmetrical in configuration to the left half, and is therefore not described separately because the person skilled in the art will immediately understand, from the detailed description of the left half, how the right half works. For example, diode  352  on the left half has a corresponding diode  352 ′ on the right half. The right-hand interlock diode  168  has the same operating principle as interlock diode  166  on the left side, and prevents MOSFETs  130  and  136  from being conductive simultaneously. 
   Signal HSR is conveyed from port RC 1  via a resistor  356  to a node  358  in logic element  160 , and signal LSR is conveyed from port RC 7  through a resistor  360  to the base of npn transistor  144 . An ALARM signal can be generated at port RC 5  if motor  100  jams, i.e. is prevented from rotating. 
   Interlock diodes  166 ,  168  serve principally to protect against uncontrollable switching states resulting from EMC-related current spikes. The switching operations (switching on and shutting off the MOSFETs) always take a certain amount of time, since gate G of the transistor in question must be charged or discharged, so that perfect protection is not possible; but this simple feature relieves a great deal of stress on the transistors in H-bridge  137  if such spikes should occur. 
   Preferred Values of Components in  FIGS. 10 AND 11   
   
     
       
             
             
             
           
             
           
             
             
             
           
         
             
                 
             
           
           
             
               Quartz oscillator 42 
               4 
               MHz 
             
             
               Capacitor 44 
               100 
               nF 
             
             
               Resistor 46 
               100 
               ohm 
             
             
               Resistors 50, 176, 302, 306, 310, 314, 320 
               10 
               kilohm 
             
             
               Capacitors 52, 308, 312, 340 
               1 
               nF 
             
             
               Hall generator 110 
               HW101G 
                 
             
             
               Op amplifiers 172, 198, 202, 300 
               LM2901P 
                 
             
             
               Resistors 134, 138 
               0.15 
               ohm 
             
             
               Resistor 178 
               75 
               kilohm 
             
             
               Resistor 182 
               33 
               kilohm 
             
             
               Resistor 186 
               15 
               kilohm 
             
             
               Transistor 188 
               BC846B 
                 
             
             
               Resistor 190 
               22 
               kilohm 
             
             
               Resistor 191 
               0.1 
               kilohm 
             
             
               Resistors 193, 309, 316 
               1 
               Megohm 
             
             
               Diodes 192, 194 
               BAV70 
                 
             
             
               Resistor 280 
               3.3 
               kilohm 
             
             
               Resistor 282 
               6.8 
               kilohm 
             
             
               Capacitor 284 
               220 
               nF 
             
             
               Resistor 297 
               2 
               kilohm 
             
             
               Resistor 304 
               1.2 
               kilohm 
             
             
               MOSFETs 114, 130, 132, 136 
               1RF7379 
                 
             
           
        
         
             
               (Component IRF 7379 contains one p-channel 
             
             
               MOSFET and one n-channel MOSFET in the same housing.) 
             
           
        
         
             
               Resistors 328, 338 
               2.2 
               kilohm 
             
             
               Diodes 330, 331, 332 
               BAW56S 
                 
             
             
               Resistors 334, 334&#39;, 344 
               5.1 
               kilohm 
             
             
               Transistors 140, 144, 148, 154 
               BC847BS 
                 
             
             
               Resistor 336 
               1.1 
               kilohm 
             
             
               Diodes 166, 168, 352, 352&#39; 
               BAS316 
                 
             
             
               Diodes 114&#39;, 118, 130&#39;, 132&#39;, 136&#39; 
               SMS2100 
                 
             
             
               Resistor 350 
               100 
               ohm 
             
             
               Resistor 346 
               330 
               ohm 
             
             
               Capacitor 348 
               4.7 
               nF 
             
             
               Resistors 342, 360 
               2.7 
               kilohm 
             
             
               Capacitor 126 
               100 
               pF, 35 V 
             
             
               Capacitor 126A 
               100 
               nF 
             
             
               Resistor 356 
               0.8 
               ohm 
             
             
                 
             
           
        
       
     
   
   These are, of course, only examples that refer here to a motor  100  which is operated on a 12-volt battery. 
   Software of Motor  100   
     FIG. 12  explains, in an overview diagram, the execution of the program steps in motor  100  as a function of the rotational position of rotor  108 . An electric motor that is controlled by a μC  40  can have a large number of additional functions depending on its application, for example rotation speed regulation, rotation speed limitation, current limitation, regulation to constant current, arrangements for outputting alarm signals, error handling routines, etc. 
   In the present exemplary embodiment, the rotation speed of the motor is regulated to a target value (e.g. 3000 rpm) that in turn can be dependent, for example, on the ambient temperature. This target value for the control program must therefore be frequently and automatically updated. 
   For a rotation speed control function, it is also necessary to know the instantaneous rotation speed of the motor, e.g. 2990 rpm. This actual value of the rotation speed also must be frequently and automatically updated. 
   It may also be necessary in such a motor to calculate acceleration; a PWM signal for the motor current must also be outputted, the calculation operations of the rotation speed control function must be performed (repeatedly), and it may be necessary to reinitialize certain parameters from time to time in order to ensure stable motor operation. 
   In addition, μC  40  must, in accordance with the calculations of the rotation speed controller, switch the current to the motor on and off, and also switch over the direction of the motor current as a function of the instantaneous rotational position. All these operations are referred to in electrical engineering as “commutation.” This should be performed with great precision, since a motor runs smoothly only if the commutations commands are executed very accurately. This means that the program must check very frequently whether a commutation program command is pending and requires execution. 
   As shown in  FIG. 12 , directly after an edge  370 ,  372  of signal HALL there is therefore a large calculation loop  374 ,  376  in which longer calculation procedures are performed depending on the value of counter HALL_CNT, followed by many short calculation loops  378  in which commutation is merely checked and, if applicable, controlled. Since these short loops  378  occur in quick succession, they result in high resolution; in other words, and as an example, every 60 to 100 μs a check is made to determine whether any changes in commutation need to be made. 
     FIG. 12  shows, for example, that directly after an edge  370  of signal HALL a long loop  374  is executed in which, as described in legend  380 , the target value for regulating the rotation speed is calculated and commutation is also checked. 
   Large loop  374  is followed by many short loops  378  in which, as shown in legend  382 , commutation is simply checked and modified as applicable. 
   In this example, an edge  372  of signal HALL is followed by a long loop  376  in which, as described in legend  384 , the following calculation steps are performed:
         Actual value calculation   Calculation of acceleration   Rotation speed regulation   Calculation of pulse duty factor pwm of signal PWM   Reinitialization of certain registers   Commutation.       

   This long loop  376  is once again followed by short loops  378  for monitoring and controlling commutation. 
   At the next edge of signal HALL, a long loop  374  of the kind already described then follows, i.e. in this exemplary embodiment, the operations repeat every 360° el. 
     FIG. 13  shows the relevant flow chart, illustrating in a rough overview the general execution of the loops just described. 
   Depicted at the very top of  FIG. 13  (at  390 ) are the interrupts, which will be described in more detail below in  FIGS. 14 through 20  and which interrupt normal program execution when they occur; this is symbolized by arrows  392 . 
   When motor  100  is switched on, an initialization of μC  40  takes place in step S 394  in the usual way. Here, in particular, a STARTUP flag is set to 1 to indicate that the program steps for accelerating motor  100  must be executed first. These steps differ from the program steps that are executed in the motor&#39;s nominal rotation speed range. 
   This is followed, in S 396 , by commutation control, which is explained in more detail in  FIGS. 23 through 26 . This control function is highly time-critical and is therefore placed at the beginning of the flow chart in a short loop  382 . 
   S 398  then checks whether the NEW_HALL flag indicates that a large loop  374  or  376  has already been cycled through since the last edge of signal HALL. 
   If this flag still has a value of 1, the program goes to S 400  where it sets the flag to 0. It then checks (in S 402 ) whether HALL_CNT is equal to either 0 or 2. (The HALL_CNT variable is generated in  FIG. 16  in S 454 . This variable corresponds to specific rotor positions that are defined arbitrarily when the motor is switched on, e.g. 0° el. and 360° el., or 180° el. and 540° el.). If Yes, the program goes into long loop  374  and, at S 404  performs the calculation of target value t_s which, in this exemplary embodiment, is calculated from the analog signal at input RA 0  (cf. FIG.  9 ). 
   If the response in S 402  is No, the program goes into long loop  376  and therein to steps S 406  and S 408 , where actual value t_HALL and the acceleration ( FIG. 29 ) are calculated. The procedure for sensing the actual value is as follows:
         Below 2000 rpm, time t_HALL is measured between two adjacent edges  370 ,  372  or  372  and  370  of signal HALL, i.e. the time to rotate through 180° el.   Above 2000 rpm, the time is measured between a first and a fourth edge of signal HALL, which in the case of the four-pole rotor  108  used here corresponds to one complete revolution of 360° mech.=720° el. In other words, the time for one complete revolution is measured, and is divided by four to obtain t_HALL.       

   These operations are explained in more detail with reference to  FIGS. 27 and 28 . 
   S 408  is followed by S 410 , where the calculation operations of rotation speed controller RGL (explained in more detail in  FIG. 30 ) are performed. 
   In S 412  that follows, pulse duty factor pwm of signal PWM is calculated, and is set at output RC 2  (cf. FIG.  31 ). 
   This is followed by S 414 , where certain registers are reset. These are registers whose values are known and do not change, e.g. registers for rotation direction or for configuration of a comparator. These registers may have lost their contents due to severe EMC-related interference. Initialization restores those contents. This is done, in the exemplary embodiment, twice per revolution of the rotor. 
   Subsequent to program steps S 404  or S 414 , the program enters an endless loop back to step S 396 . Since the NEW_HALL flag was switched over to 0 in step S 400 —meaning that one of the large loops  374 ,  376  has been cycled through—the response in S 398  is then No, and only the short loops  382  (which take a few μs) are executed. 
   At the next Hall edge  370  or  372 , the NEW_HALL flag is switched back over during the HALL interrupt to “1” (cf. S 452  in  FIG. 16 ) so that once again one of the large loops  374  or  376  is cycled through once, depending on the instantaneous value of the HALL_CNT variable. 
   If motor  100  has a four-pole rotor  108  and is rotating at 3000 rpm=50 revolutions per second, the target value and actual value are updated 100 times per second, which allows high-quality rotation speed regulation. 
     FIG. 14  shows interrupt handler S 420  that processes interrupts  390  (FIG.  13 ). The processor used here has an interrupt handler that is activated at any interrupt, identifies the interrupt in question, and then executes the necessary routine for processing that interrupt. Prior to processing of an interrupt, S 420  therefore identifies the source of the interrupt, e.g. the occurrence of a signal Imin or a change in the level of signal HALL. 
   Interrupt handler S 420  begins in S 422  by querying whether an interrupt of ring counter TIMER 1  in μC  40  is present. If so, the corresponding routine is executed in S 424 . This is part of the standard software of μC  40 . If a ring counter interrupt is not present, S 426  queries whether a Hall interrupt HALL_INT is present. If so, the corresponding routine is executed at S 428 . This is depicted in FIG.  16 . 
   If the response in S 426  is No, S 430  checks whether an Imax interrupt is present. If Yes, the Imax interrupt routine (depicted in  FIG. 17 ) is executed at S 432 . 
   If the answer in S 430  is No, S 434  checks whether an Imin interrupt is present. If Yes, the Imin interrupt routine (depicted in  FIG. 19 ) is executed in S 436 . 
   If the response in S 434  is No, S 438  checks whether a TIMEOUT interrupt is present. The TIMEOUT function has already been described in  FIG. 6 ,  242 . If such an interrupt is present, the TIMEOUT interrupt routine (depicted in  FIG. 20 ) is executed at S 440 . 
   Interrupt handler S 420  has now arrived at its end. If the response in S 438  is also No, however, then there must be an error, and the program goes to step S 442  where a corresponding error handling routine, which can be implemented in μC  40 , takes place. 
     FIG. 15  serves to explain the routine depicted in  FIG. 16  for processing a Hall interrupt. 
     FIG. 15   a  shows signal PWM at port RC 2  of μC  40 . This signal is generated continuously and has a frequency of e.g. 20 kHz. Its pulse duty factor pwm can be adjusted in program-controlled fashion (cf. FIGS.  21  and  22 ). 
     FIG. 15   b  shows signal HALL. It has leading edges  370  at the transition from Low to High, and trailing edges  372  at the transition from High to Low. 
   Times t 1 , t 2 , etc. at which the edges occur are measured by ring counter TIMER 1  and saved in a temporary variable t_TEMP. As  FIG. 15  shows, leading edges  370  govern the switching on of transistors HSL  114  and LSR  136 , i.e. of current i 1  (FIG.  1 ). Trailing edges  372  analogously govern the switching on of transistors HSR  130  and LSL  132 , i.e. of current i 2  (FIG.  1 ). The Hall interrupt routine must therefore distinguish between leading edges  370  and trailing edges  372 . 
   The time period t_HALL between two flanks is calculated as
 
 t _HALL= t   2 − t   1   (2)
 
   This duration is an indication of the instantaneous rotation speed of rotor  108 , and corresponds to the time needed by the latter to rotate 180° el. This time can, of course, be measured in many different ways, e.g. including by means of the so-called sensorless principle, using optical sensors, magnetoresistive sensors, etc. As soon as the rotation speed is high enough, it is preferable to measure the time for a larger rotation angle, in particular for one complete revolution of rotor  108 , which in the case of the exemplary embodiment according to  FIG. 1  corresponds to a rotation angle of 720° el. This measurement is explained below. 
     FIGS. 15   c  and  15   d  show, in highly schematic fashion, the signals for controlling H-bridge  137 .  FIG. 15   c  shows signals HSR, LSL for controlling transistors  130  and  132 , i.e. for switching on current i 2 .  FIG. 15   d  shows signals HSL, LSR for controlling transistors  114  and  136 , i.e. for switching on current i 1 . 
   The beginning of a pulse  444  in  FIG. 15   c  is calculated from trailing edge  372  of signal HALL, which is symbolized by an arrow  445 ; and the beginning of a pulse  446  in  FIG. 15   d  is calculated from leading edge  370  of signal HALL, as symbolized by arrow  447 . (The calculation is performed in  FIG. 30 , S 673 .) Edges  370 ,  372  of HALL correspond to defined rotational positions of rotor  108 —cf.  FIG. 26A  where a rotational position of 0° el. is associated with trailing edge  601 , a rotational position of 180° el. with leading edge  603 , etc. These are the only rotational positions that are precisely known for the calculation of commutation events, and the calculations therefore refer to these “fixed points.” 
   Assuming that control signals  444 ,  446  are located symmetrically with respect to the pulses of signal HALL, the value obtained for time t 3  at which a signal  446  begins is:
 
 t   3 = t   1 + t _HALL+( t _HALL− BW )/2  (3)
 
in which BW=the block length of signals  444 ,  446 . This block length is calculated by rotation speed controller RGL, which is described in  FIG. 30  below.
 
   The value correspondingly obtained for time t 4  at which control signal  444  should begin is:
 
 t   4 = t   2 + t _HALL+( t _HALL− BW )/2  (4).
 
   Note that time t 3 , for example, is calculated not from time t 2  (immediately preceding edge  372  of signal HALL) that is located closest to t 3 , but instead from an earlier point in time t 1 , namely from edge  370  before the previous one. The reason is that if BW=t_HALL, time t 2  would coincide with time t 3 ; this is impermissible, since calculation steps must be performed between t 2  and t 3 . 
   If a so-called ignition angle shift is used, for example by a fixed value VZ, the above formulas are modified as follows:
 
 t   3 ′= t   1 + t _HALL+(( t _HALL− BW )/2)− VZ )  (3a)
 
 t   4 ′= t   2 + t _HALL+(( t _HALL− BW )/2)− VZ )  (4a).
 
   In this case times t 3  and t 4  are located farther to the left by an amount equal to the magnitude VZ, as indicated in  FIG. 15   d  for t 3 ′; this means that currents i 1  and i 2  are switched on slightly earlier, which can result in an improvement in efficiency. It is also evident that in such a case t 3 ′ occurs earlier than t 2 , which is possible only because reference time RefTime for the calculation of t 3 ′ is not time t 2  (i.e. trailing edge  372 ) but rather time t 1  (i.e. leading Hall edge  370 ), as symbolically depicted by arrow  447 . VZ is usually a constant, but can also be a rotation-speed-dependent function or can be continuously optimized by means of separate program sections (not depicted). 
     FIG. 16  shows routine S 428  that is triggered at an edge  370 ,  372  ( FIG. 15 ) at a HALL interrupt. Such an interrupt is generated when the signal at RB 0  changes from 0 to 1 or from 1 to 0; in other words, input RB 0  is edge-sensitive and causes an interrupt upon occurrence of an edge  370  or  372 . The routine distinguishes a leading edge  370  from a trailing edge  372 , which is important for subsequent processing. 
   In step S 451 , the time at which the interrupt occurred is stored in a temporary memory t_TEMP. This point in time is measured by means of the aforementioned ring counter TIMER 1  in μC  40 . 
   In step S 452 , the NEW_HALL flag ( FIG. 13 ) is set to 1 as a signal that one of the large loops  374  or  376  ( FIG. 12 ) must subsequently be executed. 
   In step S 454 , Hall counter HALL_CNT is set to a value (HALL_CNT+1) MOD  4 , i.e. is incremented by 1 and subjected to the operation modulo  4 . The modulo calculation generates the remainder as result. For example, 4 mod 4=0, since 4 is an integer and is divisible by 4 with no remainder. 5 mod 4=1, however, since this calculation yields a remainder of 1. Similarly, 6 mod 4=2, since the remainder here is 2; 7 mod 4 is 3, and 8 mod 4=0. During operation, S 454  therefore continuously yields the number sequence 0, 1, 2, 3, 0, 1, 2, 3, 0 etc. for HALL_CNT. 
   Step S 456  queries whether HALL=HIGH. According to  FIG. 12   a ), this means that rotor  108  is in an angular position between 0° el. and 180° el. 
   If HALL is not high, then in S 458  the reference variable for controlling high-side right transistor HSR  130  and low-side left transistor LSL  132  is replaced by the time stored in temporary memory t_TEMP. In the next step S 460 , the interrupt sensitivity is set so that port RB 0  is sensitized, for the next HALL interrupt, to a change from LOW to HIGH. 
   S 462  checks whether the COMMUT_ON flag has a value of 0. This flag is set in the COMMUT routine ( FIG. 23 ) in step S 718  as soon as the winding receives current, and is set to zero at the completion of commutation in  FIG. 24  or  25  (cf. S 764 , S 812 , and S 842  therein). If the response is No, this means that a current i 2  is still flowing at the time of the Hall change from High to Low. 
   Referring again to  FIG. 15 , a Hall change  372  from High to Low occurs therein at time t 2 . Transistor HSR  130  should already have been shut off there so that current i 2  no longer flows, and since HSR is still conductive, i 2  must be shut off in an “emergency shutoff” procedure. To achieve this, in step S 464  HSR  130  is shut off, and in the next step S 466  both low-side transistors LSL  132  and LSR  136  are switched on, so that current i 2  can decay rapidly through transistors  132 ,  136  and measuring resistors  134 ,  138  and thereby generate a torque. (When current i 2  transitions through zero, an Imin interrupt according to  FIG. 19  is triggered, terminating the shutoff procedure.) The program then goes to step S 468 , where it is now determined that the shutoff procedure for current i 2  has been initiated (COMMUT_ON:=0), which according to  FIG. 23 , S 702  is the prerequisite for switching on current i 1 . 
   If it is found in step S 462  that current i 2  has already been shut off, the program goes directly to step S 468 . 
   If it is found in step S 456  that signal HALL is high, i.e. that the edge in  FIG. 15  is a leading edge  370 , the program goes to step S 470 , where the time stored in temporary memory t_TEMP is taken as the reference variable for controlling transistors HSL  114  and LSR  136 , i.e. certain times are now measured and calculated from that variable. In S 472  the interrupt sensitivity is then set so that port RB 0  is sensitized, for the next HALL interrupt, to a change from HIGH to LOW, i.e. to a trailing edge. 
   The next step S 474  checks whether the COMMUT_ON flag has a value of 0. This flag is set to 1, in the COMMUT routine ( FIG. 23 ) in step S 718 , as soon as the winding receives current, and is set to zero at the completion of commutation in  FIG. 24  or  25  (cf. S 764 , S 812 , and S 842  therein). If the response is No, because a current i 1  is still flowing at this Hall change, that current must be shut off in an “emergency shutoff” procedure, for which purpose current i 1  is shut off in step S 476  by shutting off high-side transistor HSL  114 , and in S 478  both low-side transistors LSL  132  and LSR  136  are switched on, so that current i 1  can decay rapidly through components  132 ,  134 ,  136 ,  138  and thereby generate a torque on rotor  108 . (When current i 1  transitions through zero, the shutoff procedure is terminated, e.g. by means of the Imin interrupt of  FIG. 19. ) S 468  then follows, in which COMMUT_ON is set to 0 in order to indicate that the shutoff procedure for i 1  has been initiated. If the response in S 474  is Yes, the program goes directly to step S 468 . 
   Following S 468 , S 480  checks whether the STARTUP flag ( FIG. 13 , S 394 ) has a value of 1. This means either that no value at all is present for the actual rotation speed, or that the actual rotation speed is less than 1000 rpm. If this flag is not set, the program branches directly to the end S 493  of routine S 428 . 
   If the response is Yes in S 480 , the program goes to step S 482  and checks there whether t_HALL is less than a value t_HALL_min (cf. equation (7)) which value corresponds e.g. to a rotation speed of 1000 rpm, i.e. it determines whether the rotation speed has risen above 1000 rpm. If No, the program goes to S 493 . 
   If the rotation speed has risen above 1000 rpm, the STARTUP flag is set to zero in S 486 . S 488  then checks whether signal HALL is high. If No, S 490  defines in the NEXT_COMM predictive variable that the next current block will be a current block  446  (FIG.  15 ), i.e. that HSL  114  and LSR  136  must be switched on in it. If the response in S 488  is Yes, it is then stipulated at S 492  that the next current block will be a current block  444  (FIG.  15 ), i.e. that HSR  130  and LSL  132  must be switched on in it. After S 490  or S 492 , the program goes to S 493  and terminates routine S 428 . The values for NEXT_COMMUT are queried in  FIG. 24 , S 752  and  FIG. 25 , S 806 , and enable the transition to commutation at higher rotation speeds. 
     FIG. 17  shows a preferred embodiment of routine S 428  for processing an Imax interrupt S 428 . The operation of this routine is then explained with reference to FIG.  18 . 
   Step S 500  checks whether the Imax_CTRL_ON flag was set to 1 in the COMMUT_CTRL routine (FIG.  25 ). The result of this is that routine S 428  can be initiated by signal Imax only if a current is flowing in winding  102 , but not by interference signals when the winding is currentless. If the response in S 500  is Yes, S 501  checks whether the Imax_Interrupt was generated at the upper limit (3 A) or lower limit (1.6 A). For an interrupt at the upper current limit, signal Imax goes from High to Low because transistor  188  ( FIG. 1 ) becomes conductive, and the current to stator winding  102  has already been shut off by the hardware by means of the low-active signal Imax, by blocking both high-side transistors  114  and  130 . This has already been described in FIG.  1 . Additionally and redundantly, if the response in S 501  is Yes, in S 502  signals HSL_OUT and HSR_OUT for highside transistors  114  and  130  are set to zero for additional control of these two transistors by means of software, i.e. they can be switched back on only when permitted by the software. If the response in S 500  is No, the routine goes directly to its end, i.e. S 522 . The routine also goes directly to S 522  if the interrupt was generated at the lower current limit (1.6 A) (S 501 : No). 
   At S 504 , S 502  is followed by a 30-μs wait time. During this time, the current in the lower portion of bridge  137  flows, for example, through conducting transistor  136  and free-wheeling diode  132 ′, or conversely through conducting transistor  132  and free-wheeling diode  136 ′. 
   Then, at S 506 , both low-side transistors LSL  132  and LSR  136  are switched on so that the current in winding  102  can decay through components  132 ,  134 ,  136 ,  138 , generating a torque on rotor  108 . 
   Next, at S 508 , comes the DEC*(pwm) routine, which is depicted in FIG.  22  and in which pulse duty factor pwm of signal PWM is reduced one step so that the current through winding  102  decreases and no longer reaches the upper limit (here 3 A). The result is to adaptively prevent the motor from operating unnecessarily with current limiting, and the reduced current is compensated for by increasing the value BW (in controller RGL). 
   This is followed in step S 510  by a wait time of e.g. 200 μs so that the current in winding  102  has enough time to decay. S 511  checks whether the variable for the next transistors to be switched on is HSL/LSR. If Yes, then in S 512  transistor LSR  136  remains conductive and transistor LSL  132  is shut off, so that the short-circuit current now flows through transistor  136  and free-wheeling diode  132 ′. Following this in S 512  there is a wait time of e.g. 30 μs, and then high-side transistor HSL  114  is once again made ready to be switched on, i.e. it can be switched on by the hardware when signal Imax becomes high. This is indicated symbolically in  FIG. 17  at  513  by “Hardware: ON.” Switching on is therefore accomplished not by means of the command HSL_OUT:=1, but only by a logical association between this signal and the change in signal Imax when the current drops below 1.6 A. Below 1.6 A the motor therefore immediately begins receiving energy from DC power network  121  again, and i 1  rises again. 
   If the response in S 511  is No, then in S 514  transistor LSR  136  is blocked and transistor LSL  134  remains switched on (cf. S 506 ), so that the short-circuit current flows through transistor  134  and free-wheeling diode  136 ′. There is then a 30 μs wait time, and high-side transistor HSR  130  is then once again made ready to be switched on, i.e. it can now be switched on by the hardware, as indicated at  513 , as soon as signal Imax once again becomes high, i.e. at a current below 1.6 A. Here again, switching on is accomplished not by means of signal HSR_OUT:=1, but only by way of the change in signal Imax at 1.6 A, in other words by means of a conjunctive association between signal HSR_OUT:=1 and signal Imax =1. After the current drops below 1.6 A, motor  100  therefore once again receives current from DC link  121 , and current i 2  rises again. 
   Subsequent to S 512  or S 514 , routine S 428  goes to S 522  where it ends. 
   It should be noted here that signals HSL_OUT, HSR_OUT, etc. remain stored until a different signal is generated at the relevant output of μC  40 . Subsequent to S 502 , for example, signal HSL_OUT remains at 0 until it is switched over to 1 in S 512 , and subsequent to S 512  it remains at 1 until it is switched back to 0 at some other time. 
     FIG. 18  explains the manner of operation of the routine shown in FIG.  17 . In  FIG. 18 , a value of 3 A is shown for the upper current threshold, and a value of 1.6 A for the lower current threshold, in order to improve comprehension. These numerical values may, of course, be different depending on the motor. 
   At t 30 , current i 1  is switched on by switching on transistors  114  and 136. At t 31 , i 1  reaches the permissible maximum value of 3 A, and as a result of the change in signal Imax to Low, transistor  114  is immediately shut off by the hardware. At the same time, starting at t 31 , routine S 428  is executed as shown in FIG.  17 . This routine, by means of S 506 , additionally switches on low-side transistor  132  at t 32 , so that winding  102  is operated in short circuit. This lasts for 200 μs until t 33 , when transistor  132  is shut off again so that only transistor  136  is conductive, and the software shutoff of high-side transistor  114  is cancelled by steps S 516  and S 518 . High-side transistor  114  does not conduct until after t 34 , however, namely when the lower current threshold of 1.6 A is reached, thereby making signal Imax high again so that current i 1  is switched on and rises again. At t 35  it again reaches the 3 A level, and transistor  114  is once again shut off by the hardware, routine S 428  is started again, and the procedure just described repeats. 
   At t 36  transistor  114  is once again switched on by the hardware, and at t 37  the shutoff command becomes effective because the duration BW of the current block has elapsed. 
   Inherently, current i 1  should already have been shut off at point Z at which time BW elapsed, but the shutoff command can take effect only in the areas shaded in gray in  FIG. 18 , i.e. in this case not until time t 37 , with the result that the shutoff is slightly delayed. 
   At time t 38  current i 1  transitions through zero, and Imin interrupt S 436  (described below) is therefore generated there. 
   It is somewhat disadvantageous in the context of  FIGS. 17 and 18  that increased losses occur, for example, between times t 33  and t 34  because i 1  is then flowing through free-wheeling diode  132 ′ because transistor  132  is no longer conductive. A variant with which these losses can be further reduced, and which is especially suitable for slow motors, will also be described below. The approach according to  FIGS. 17 and 18  represents the optimum for fast-running motors based on present knowledge, since in such motors the current changes occur extremely fast and therefore the calculation times in μC  40  are too long compared to the times within which those current changes take place. Even better solutions would probably be possible with faster processors, but at present these are still too expensive for motors. 
     FIG. 19  shows the execution of service routine  436  for processing an Imin interrupt. 
   S 530  queries whether the Imin_INT_ON flag is equal to 1. This flag is set in the COMMUT_CTRL routine ( FIG. 25 , S 824 ). If a TIMEOUT interrupt ( FIG. 20 ) has directly preceded (FIG.  20 ), this flag has a value of 0, and the program goes directly to the end, i.e. to S 532  of this routine. 
   If the response in S 530  is Yes, in S 534  the TIMEOUT_INT_ON flag is set to 0 so that a subsequent TIMEOUT interrupt is no longer processed; and then at S 536  all four transistors  114 ,  130 ,  132 ,  136  are blocked, because winding  102  is substantially currentless and contains no stored inductive energy (which has been converted into kinetic energy of rotor  108 ). 
   Then, at S 538 , the BlockEnd_DONE flag (which is queried in  FIG. 24  at S 762  during the COMMUT_NORMAL routine and serves to prepare the next commutation) is set to 1, and at S 539  Imin_INT_ON is set to 0 because the routine has been executed. 
     FIG. 20  shows the execution of service routine S 440  for processing a TIMEOUT interrupt. 
   S 540  queries whether the TIMEOUT_INT_ON flag has a value of 1. If an Imin interrupt ( FIG. 19 ) has preceded, this flag has a value of 0, and in that case the routine goes directly to its end at S 542 . 
   If the response in S 540  is Yes, the routine goes to step S 544  where it sets the Imin_INT_ON flag to 0 so that a subsequent Imin interrupt is not processed (cf. S 530  in FIG.  19 ). 
   In the next step (S 546 ), all four transistors  114 ,  130 ,  132 ,  136  are blocked because the current in winding  102  has a low value at the expiration of TIMEOUT, and winding  102  is consequently no longer storing a large amount of inductive energy. Winding  102  is thereby made currentless. 
   At S 548  the BlockEnd_DONE flag (which is queried in  FIG. 24  in S 762  during the COMMUT_NORMAL routine) is then set to 1, and in S 549  TIMEOUT_INT is set to 0 because the interrupt has been processed. 
     FIG. 21  shows INC*(PWM) routine S 554  for increasing pulse duty factor pwm of signal PWM at output RC 7  of μC  40 . At S 556  the value in the PWM register is incremented by 1, corresponding to a 1% increase in the pulse duty factor. 
   Step S 558  checks whether the increase has caused pwm to become greater than 100%. If Yes, the program goes to S 560 , where pwm is then set to 100%, meaning that current i 1  or i 2  is switched completely on. 
   If the response in S 558  is No, the routine goes to its end S 562 ; the same occurs subsequent to S 560 . 
     FIG. 22  shows DEC*(PWM) routine S 564  for decreasing pulse duty factor pwm. At S 566  the pwm variable is decremented by 1, corresponding to 0.5%. S 568  checks whether this has caused pwm to drop below 10%. If Yes, the routine goes to S 570  where a lower limit of 10% is imposed on pwm. If the response in S 568  is No, the routine goes to its end S 572 ; the same occurs subsequent to S 570 . 
   The routines shown in  FIGS. 21 and 22  play a part principally in the context of the adaptive controller, which is described below with reference to FIG.  31 . 
     FIGS. 23 through 25  shown COMMUT routine S 396 , which is continuously called in the main program ( FIG. 13 ) and controls currents i 1 , i 2 , in winding  102 . Commutation control is the function executed most frequently. It comprises two sections: 
   1. The start-up section for starting and acceleration; 
   2. The section for normal operation. 
   In the start-up program section, the motor is at a standstill or is just beginning to accelerate. Once supply voltage has been connected, the STARTUP flag is set in  FIG. 13  in step S 394  so that the motor begins the STARTUP routine. The COMMUT_ON flag is also set to 0 during initialization so that a new current flow operation can start. 
   S 700  checks whether motor  100  is in start-up (STARTUP=1). If Yes, execution branches to S 702  and a simplified commutation is performed. 
   Commutation at Low Rotation Speeds 
   At low rotation speeds, the current through winding  102  is switched on by means of COMMUT routine S 396  (FIG.  23 ), and is shut off again in the respectively subsequent Hall interrupt routine (FIG.  16 ). S 702  first checks whether the current block has already been started in this Hall period. If Yes, execution branches to the end S 722 , since a current flow will take place only after the next Hall change. If it was found in S 702  that COMMUT_ON=0, however, this is the first call of COMMUT routine S 396 , and current flow is started. 
   To accomplish this, there is a 100 μs wait time in S 704  to create a current gap so that the MOSFETs are not all conductive simultaneously. S 706  checks whether block length BW is greater than zero. If No, the motor should receive no current. The routine therefore branches to the end S 722 . 
   If BW&gt;0, the correct current flow to winding  102  (i.e. either i 1  or i 2 ) is started as a function of signal HALL (cf. FIG.  1 ). Rotor  108  then begins to rotate 180° el. 
   If HALL is high, signals HSR_OUT and LSL_OUT are then set to 1 in S 710  so that winding  102  experiences current flow through transistors HSR  130  and LSL  132 , and a current i 2  flows. 
   S 712  defines predictively that the next commutation must occur via transistors HSL  114  and LSR  136 . This is important for changing from this commutation mode to the commutation mode at high rotation speed (cf. the description of  FIG. 27 , below). 
   If, however, signal HALL was found in S 708  to be Low, then in S 714  the other transistors HSL  114  and LSR  136  are switched on so that a current i 1  flows; and in S 716  NEXT_COMMUT is predictively set to the correct value for the next commutation. 
   Lastly, in S 718  the COMMUT_ON flag is set to 1 so that at the next call of COMMUT routine S 396 , execution branches directly from S 702  to S 722 , since winding  102  is already receiving current. This continues until rotor  108  has rotated approximately 180° el. 
   Once 180° el. has been reached, the software detects this by way of a Hall interrupt. The shutoff of current flow, and the setting of COMMUT_ON to 0, are performed in the Hall interrupt routine ( FIG. 16 , S 462  through  478 , S 468 ), so that the commutation control function once again, beginning at S 704 , starts a new current flow in the correct current direction. 
   Commutation at High Rotation Speeds 
   If STARTUP=0 in S 700 , the COMMUT_NORMAL commutation routine S 720  for high rotation speeds is performed (cf. FIG.  24 ).  FIG. 26  shows a schematic diagram illustrating the execution of this commutation function. 
   In S 750  in  FIG. 24 , the instantaneous time t_TIMER 1 , which is continuously measured by a ring counter, is stored in the t_CALC variable; and in S 752  a decision is made, based on the NEXT_COMMUT variable, as to the direction in which current is to flow through winding  102 . 
   If transistors HSL and LSR are to be switched on, execution branches to S 754  and the RefTime_HSL/LSR variable, which corresponds to the time of the previous Hall change from Low to High, is subtracted from the t_CALC variable. This is depicted in FIG.  26 .  FIG. 26A  shows signal HALL with Hall changes  601 ,  603 ,  605 ,  607 , etc. during which the time of the instantaneous Hall change is stored (S 458  and S 470  in  FIG. 16 ) in the respective variables RefTime_HSR/LSL (at  601  and  605 ) and RefTime_HSL/LSR (at  603  and  607 ). 
     FIG. 26  explains the basic principle of commutation. For switching on and shutting off a current block, reference is made, after the motor has reached operating speed, to a reference position of the rotor, associated with that current block, which maintains a minimum distance from that current block in all operating states. 
   For example, a reference position ∂ 0  (here 180° el.) is used for switching on and shutting off current block B 4  (FIG.  26 C), and from that reference position ∂ 0  an angular position ∂ 1  is calculated for switching on current block B 4  (in this case at 405° el.), as well as an angular position ∂ 2  for shutting off block B 4  (in this case at 495° el.). 
   Angular position ∂ 0  is therefore the reference point for this current block, and a reference time RefTime_HSL/LSR is therefore measured in TIMER 1  at that position, since transistors HSL  114  and LSR  136  must be conductive in current block B 4 . 
   Motor  100  does not have a sensor with which rotation angle ∂ could be exactly measured in every case; instead, the rotational position can be sensed with some accuracy only at four positions where signal HALL changes, namely at 0° el., 180° el., 360° el., and 540° el. Interpolation is required between these rotational positions; this is possible because there is little change in the angular velocity of rotor  108  in the course of one revolution. 
   If the intention is therefore to switch on at rotational position ∂ 1  and shut off at position ∂ 2 , it is known that the angular distance between ∂ 0  and ∂ 1  is, for example, 405−180=225° el., and that the angular distance between ∂ 0  and ∂ 2  is, for example, 495−180=315° el. 
   Since it is known that the rotor requires a time t_HALL to rotate 180° el., the time resulting for a rotation of 225° el. is
 
 t _HALL*(225/180)=1.25* t _HALL
 
In this example, this is the time t_BLOCK_START.
 
   The time obtained for 315° el. is similarly
 
 t _HALL*(315/180)=1.75* t _HALL
 
In this example, this is the time t_BLOCK_END.
 
   When rotational position ∂ 0  is passed through, a reference time is therefore measured, i.e. RefTime_HSL/LSR, e.g. 67.34 ms. 
     FIG. 33  shows the values indicated above in a quantitative example for n=3000 rpm. According to equation (6), time t_HALL=5 ms. This is the time required for rotor  108  to travel 180° el. at 3000 rpm. 
   Controller RGL ( FIG. 30 ) specifies at  613  (as an example) a block length BW of 2.5 ms, and it is therefore known predictively from  FIG. 33  that rotational position ∂ 1  (405° el.), at which current i 1  must be switched on, will be reached after a period of 6.25 ms. It is also known predictively that rotational position ∂ 2  (495° el.) at which current i 1  must be shut off and at which the commutation procedure begins and energy delivery from the DC link must be shut off, will be reached after a period of 8.75 ms. 
     FIG. 33  furthermore shows, as an example, that a reference time of 65.34 ms is measured in TIMER 1  at reference time ∂ 0 . This is the time RefTime_HSL/LSR. 
   The procedure for monitoring switching on at ∂ 1  and shutoff at ∂ 2  is, as shown in  FIG. 24  at S 754 , to continuously calculate the time difference t_CALC between 65.34 ms and the instantaneously measured time (cf. equation (5) regarding t_CALC). 
   If a time of 66.34 ms is measured, for example, at time t 40  by TIMER 1 , the resulting difference is then
 
 t _CALC=66.34−65.34=1 ms.
 
Since current i 1  needs to be switched on only after a period of 6.25 ms, 1 ms is not long enough and current i 1  is not yet switched on.
 
   If the present time in TIMER 1  at time t 41  is 71.60 ms, the resulting difference is then
 
 t _CALC=71.60−65.34=6.26 ms.
 
In this case current i 1  is switched on, since t_CALC is greater than 6.25 ms.
 
   Starting at rotational position ∂ 0 , therefore, there is constant monitoring (in  FIG. 25 , S 800 ) as to whether t_CALC has become greater than t_BLOCK_START; and if that is the case, transistors HSL  114  and LSR  136  are then switched on in this case in S 810  of FIG.  25 . 
   Shutoff is accomplished on the same principle, except that t_CALC is compared to the t_BLOCK_END variable (cf. S 820  in FIG.  25 ). In  FIG. 33  this variable is 8.75 ms. It corresponds to shutoff angle ∂ 2 , and when it is reached, the commutation procedure according to  FIG. 25 , S 826  through S 844  is executed. 
   Commutation is therefore based on recalculating time t_CALC in the short loops  382  of  FIG. 13 , at very short intervals of e.g. 0.1 ms, and comparing it to the predictive values t_BLOCK_START and t_BLOCK_END. This occurs in  FIG. 33A  between the times 65.34 ms and 74.1 ms, and is indicated by dots  615 . The departure point for each current block is a reference angle, associated with that block, at which a reference time is measured for that current block and is then used in the comparisons. As rotor  108  rotates, new reference times are continuously being determined and new comparisons made, so that currents i 1  and i 2  through winding  102  are correctly controlled, i.e. the reference angles continuously “migrate” as the rotor rotates. The same principle can of course also be applied to motors having more than one winding. 
   If the current is to be switched on earlier, by an amount equal to a time ZV=0.4 ms (also referred to as “ignition advance”), what is then used in  FIG. 33  instead of the 6.25 ms time for switch-on is a time of
 
6.25−0.4=5.85 ms,
 
and for shutoff:
 
8.75−0.4=8.35 ms.
 
   At this rotation speed, angle ∂ 1  then shifts 14.4° el. to the left to 390.6° el., and angle ∂ 2  also shifts, at this rotation speed, 14.4° el. Toe the left to 480.6° el., i.e. current i 1  is switched on and shut off at earlier times, and the angle defining how much earlier is it switched on and shut off increases as the rotation speed rises; in this case (at 3000 rpm), it is 14.4° el., 28.80 el. at 6000 rpm, etc. ZV will usually be a function of rotation speed. This earlier switching on of the currents in winding  102  can improve the efficiency of motor  100  at higher rotation speeds. It is very easy to implement with the present invention. 
     FIG. 26B  shows the value of the NEXT_COMMUT variable, i.e. either HSL/LSR or HSR/LSL.  FIG. 26C  symbolically shows current-flow blocks B 1  through B 5  plotted against time TIMER 1 .  FIG. 26D  shows times t_BLOCK_START and t_BLOCK_END for current-flow block B 4 , which begins at  609  and ends at  611 . Block B 4  has, as reference time for being switched on and shut off, edge  603  of signal HALL, i.e. the time RefTime_HSL/LSR( 603 ) measured at  603 , which is symbolized in  FIG. 26C  by an arrow  611 . At  621 ,  623 ,  625 , and  627  the time span
   t _CALC= t _TIMER 1 −RefTime_HSL/LSR  (5) 
is adapted (by recalculation in the program) to the present time in TIMER 1 . For example, at  621  a time t_CALC( 621 ′) is calculated for time  621 ′, and is used to check whether the beginning of block B 4  has already been reached.
 
   At times  621 ′,  623 ′,  625 ′, and  627 ′, the NEXT_COMMUT variable ( FIG. 26B ) has the value HSL/LSR, so that execution branches from S 752  ( FIG. 24 ) to S 754 , where the instantaneous difference between the value t_TIMER 1  stored in S 750  and the value RefTime_HSL/LSR( 603 ) is calculated and is assigned to the t_CALC variable. When the COMMUT_NORMAL routine is called at time  621 ′, the t_CALC variable therefore has the value indicated at  621  (FIG.  26 D). The analogous calculation takes place at S 756  if the NEXT_COMMUT variable has the value HSR/LSL. 
   Execution thereupon branches into the actual commutation routine COMMUT_CTRL S 760 , which is depicted in FIG.  25 . The portion of  FIG. 24  beginning at S 762  serves to terminate commutation, i.e. Toe shut off the current; it is executed only after the completion of current flow and will be described later. 
   In COMMUT_CTRL routine S 760 , transistors  114 ,  130 ,  132 , and  136  are switched on and shut off, as described with reference to FIG.  26 . 
   If the time span calculated in t_CALC (e.g. at time  621 ′) is less than t_BLOCK_START, no current flow should take place through winding  102 . 
   At  623 ′, t_CALC is for the first time greater than t_BLOCK_START, and the current to winding  102  is therefore switched on. 
   At time  625 ′, the value t_CALC has not yet reached the value t_BLOCK_END, so current flow through winding  102  is continued. 
   At  627 ′, t_CALC has finally exceeded the time span t_BLOCK_END, and energy delivery to winding  102  is therefore now shut off. 
   The steps just recited are performed in COMMUT_CTRL routine S 760 . If t_CALC in S 800  is less than t_BLOCK_START (time  621 ′), then nothing happens and execution branches to the end S 848 . 
   If, however, t_CALC in S 800  is greater than or equal to t_BLOCK_START (times  623 ′,  625 ′,  627 ′), S 802  then checks whether current flow to winding  102  is already activated (COMMUT_ON=1). If No (time  623 ′), the switch-on procedure takes place starting at S 804 . 
   If block length BW=0 in S 804 , then no current is delivered and execution branches to S 812 . If, however, BW&gt;0, then depending on the value of the NEXT_COMMUT variable, transistors HSR  130  and LSL  132  are made conductive in S 808 , or transistors HSL  114  and LSR  136  in S 810 . 
   In S 812 , COMMUT_ON is set to 1 to indicate that current flow to winding  102  is now switched on. Execution then branches to the end S 848 . 
   If the value COMMUT_ON=1 in S 802  (times  625 ′,  627 ′), i.e. if a current is flowing to winding  102 , S 820  then checks whether the t_CALC variable has already reached the value of time span t_BLOCK_END that is calculated in  FIG. 30 , S 673 . 
   If No (time  625 ′), S 822  additionally checks whether t_CALC is greater than or equal to (2*t_HALL−A*). For this motor, (2*t_HALL) is the time needed for rotor  108  to rotate 360° el., and A* is a constant equal to, for example, 400 μs. The effect of S 822  is to interrupt current to the winding approximately 400 μs before the next Hall change, even in the event of disruptions in program execution. 
   That 400-μs period is needed so that the entire shutoff procedure can be executed before the Hall change occurs. The purpose of this is to prevent simultaneous activation of all the power transistors. This “emergency shutoff” is necessary at high rotation speeds because at such speeds, block length BW is almost as great as t_HALL (high power requirement at high rotation speed). At low rotation speeds, the end of a current block is already reached long before the next Hall change occurs, i.e. the response in S 822  is always No, and in S 824  the Imax interrupt ( FIG. 17 ) is activated to allow reaction, if necessary, to an excessive motor current. 
   If, however, the value of t_CALC in S 820  is greater than or equal to t_BLOCK_END, or if the response in S 822  is Yes, the shutoff procedure is then called in S 826 . 
   S 826  checks, on the basis of the Off_detected variable, whether the shutoff of current flow (i.e. the commutation procedure for shutoff) has already been initiated. If Yes, execution branches to the end S 848 . If this is the first call, however, execution branches from S 826  to S 828 . 
   The Off_detected variable is set to 1 in S 828 . In S 830  the Imax interrupt is deactivated, and in S 832  the Imin interrupt is activated. (It is very advantageous if the interrupts are activated only in the regions in which they can occur in accordance with the program&#39;s logic.) 
   In S 834  both high-side transistors HSL  114  and HSR  130  are shut off. In S 836  there is a 30-μs wait time, and in S 838  the TIMEOUT interrupt ( FIG. 20 ) is activated and a TIMEOUT time t_TIMEOUT is calculated from the instantaneous value of TIMER 1  and a constant t_T 0 . 
   At S 840 , both low-side transistors LSL  132  and LSR  136  are then made conductive so that the current in winding  102  decays in short circuit and can thereby generate kinetic energy in rotor  108 . At S 842  the COMMUT_ON flag is set to 0, and in S 844  the BlockEnd_DONE variable is set to 0 to indicate that commutation is not yet completely finished. Whichever of the two interrupt routines (Imin interrupt and TIMEOUT interrupt) is called first then shuts off both low-side transistors LSL  132  and LSR  136  (cf. S 536  of FIG.  19  and S 546  of  FIG. 20 ) and sets BlockEnd_DONE to 1 (cf. S 538  of FIG.  19  and S 548  of FIG.  20 ). Shutoff is thereby completely terminated, and this is indicated by BlockEnd_DONE=1. 
   At the next call of COMMUT_NORMAL routine S 720 , in  FIG. 24 , S 762  execution branches to S 764 . In S 764 , COMMUT_ON and Off_detected are set to 0 because current flow is shut off, and in S 766  through S 770  the predictive value of NEXT_COMMUT is changed, i.e. the value HSL/LSR is changed to LSR/LSL and vice versa (cf. FIG.  26 B). The result is that even in the context of an “ignition advance,” in which the current is switched on before the actual associated Hall interrupt, the direction of current flow in winding  102  is defined correctly, i.e. the value of NEXT_COMMUT defines which transistor pair needs to be monitored next in terms of switching on and shutting off. In S 772  the BlockEnd_DONE flag is then set to 0 so that at the next pass the response in S 762  is No, and the routine branches directly to S 774 . 
     FIG. 27  shows CALC_t_HALL routine S 406  for calculating the instantaneous Hall time t_HALL, i.e. the time needed for rotor  108  to rotate through 180° el. 
     FIG. 28  is an overview for explanatory purposes.  FIG. 28D  shows signal HALL, which has edges at points  630 ,  631 ,  632 ,  633 ,  634 ,  635 , at each of which a Hall change occurs that is used to determine the rotor position and to determine rotation speed and acceleration. Since a Hall change takes place four times per revolution with a four-pole rotor  108 , the exact rotor position is measurable four times per revolution. 
     FIG. 28B  shows the value of the HALL_CNT variable. This is a counter which (according to S 454 ,  FIG. 16 ) is incremented modulo  4 . This means that this variable sequentially assumes the values 0, 1, 2, 3, 0, 1, 2, 3, 0 . . . 
     FIG. 28A  shows, by way of example the position of rotor  108 , which is depicted as a four-pole rotor as in FIG.  1 . Edge  630  of signal HALL corresponds to the 0° el. rotor position and to counter status HALL_CNT=0, edge  631  to the 180° el. rotor position and to counter status HALL_CNT=1, edge  632  to the 360° el. rotor position and to counter status HALL_CNT=2, etc. 
   Two measurement approaches are used.  FIG. 28E  shows the one approach which is used at low rotation speeds n, e.g. at less than 2000 rpm, where t_HALL assumes large values (cf. equations (6) and (7) below).  FIG. 28F  shows the other approach which is used at higher rotation speeds (e.g. above 2000 rpm), at which Hall times t_HALL are shorter and inaccuracies due to magnetization defects of rotor  108  are avoided by measuring the time for one complete revolution (720° el.). 
   CALC_t_HALL routine S 406  is called by the main program ( FIG. 13 ) at every second Hall interrupt, specifically when the HALL_CNT variable ( FIG. 28 ) is an even number, i.e. has a value of either 0 or 2 (cf. step S 402  in FIG.  13 ). 
   The instantaneous time of the Hall change was previously stored in Hall interrupt routine S 428  (FIG.  16 ), specifically in RefTime_HSR/LSL for an edge from High to Low (S 458  in  FIG. 16 ;  FIG. 28C ) and in RefTime_HSL/LSR for an edge from Low to High (S 470  in  FIG. 16 ; FIG.  28 C). At rotor positions 0° el., 360° el., 720° el., etc., the time for the relevant rotor position is therefore stored as a reference time for HSL/LSR, and at rotor positions 180° el., 540° el., 900° el., etc. the time for the relevant rotor position is stored as a reference time for HSR/LSL, as indicated explicitly in FIG.  28 C. 
   In S 851  or S 852  (FIG.  27 ), depending on the value of signal HALL, the time span between the instantaneous and previous Hall change is calculated and is stored in the TEMP variable. In  FIG. 28E , for example after Hall change  632 , this would be the time between edges  631  and  632 , i.e. [RefTime_HSL/LSR ( 632 )—RefTime_HSR/LSL ( 631 )]. In S 854  the instantaneous time t_HALL is stored in t_HALL_OLD so that an acceleration calculation can be performed (cf. FIG.  29 ). 
   S 856  checks whether the time span TEMP is shorter than the time span t — 2000 (time t — 2000 being equal to time t_HALL at 2000 rpm). If No, rotation speed n of motor  100  is less than 2000 rpm, and the left branch S 858 , S 860  is executed, in which time t_HALL is calculated for one-quarter of a revolution, i.e. for 180° el. In S 858 , the value TEMP from S 851  or S 852  is assigned to Hall time t_HALL, and in S 860  FLAG — ¼ is set to 1 to indicate that at present, only the time for a quarter-revolution is being measured. 
   If it is found in S 856  that the rotation speed of the motor has already reached a rotation speed n=2000 rpm, S 862  then checks whether the HALL_CNT variable equals 0. This is true after each complete mechanical revolution of rotor  108  (cf. FIGS.  28 A and  28 B). If No, execution branches immediately to the end S 878 , e.g. in the case of edge  632  in FIG.  28 D. If, however, HALL_CNT=0, S 864  then checks whether FLAG — ¼=1. 
   If Yes, this is the very first pass through the t_HALL calculation for one complete rotor revolution, and therefore for this pass the present value RefTime_HSL/LSR is stored in RefOld so that starting with the next pass, it is possible to calculate using a valid value for RefOld. At the very first pass, there is no calculation of t_HALL over one complete mechanical revolution, but instead the previous value is re-used. In S 866  FLAG — ¼ is set to zero, i.e. starting with the next pass the measurement can be made over one complete revolution of rotor  108 . 
   At the next call of CALC_t_HALL S 406 , at which HALL_CNT=0, execution branches from S 864  to S 868 . There the time span is calculated between the instantaneous value RefTime_HSL/LSR (e.g. from edge  634  of  FIG. 28D ) and the value stored one rotor revolution ago in RefOld (e.g. at edge  630  of FIG.  28 D). This time span corresponds to four times the Hall time t_HALL, and in S 870  the calculated value is therefore divided by four so that the value t_HALL corresponds to exactly one-quarter of the time required for one entire revolution (from  630  to  634  in  FIG. 28E , i.e. 720° el.). This approach to measuring t_HALL is particularly accurate, and therefore results in particularly smooth motor operation. 
   In S 874  the instantaneous value RefTime_HSL/LSR for the next calculation is stored in the RefOld variable. Execution then leaves the routine in S 878 . 
   The time RefTime_HSR/LSL could similarly be used instead of RefTime_HSL/LSR, as is self-evident to one skilled in the art. The choice depends on the rotor position at which counter HALL_CNT has a counter status of 0. 
   In this exemplary embodiment, CALC_t_HALL routine S 406  is called only after every second Hall interrupt because of branch S 402  in the main program (FIG.  13 ). The query in S 402  of  FIG. 13  ensures that when it is called, it has available to it the correct reference times for rotation speed calculation over one complete revolution. 
   With a fast processor, that same CALC_t_HALL routine S 406  could also be called more frequently. 
     FIG. 29  shows CALC_ACCEL routine S 408  which is used to calculate the acceleration of rotor  108 . As shown in  FIG. 13 , this routine is executed subsequent to the CALC_t_HALL routine which prepares (in step S 854 ) for the execution of routine S 408 . 
   In step S 640 , the ACCEL variable is calculated as the difference between t_HALL_OLD and t_HALL. 
   S 642  checks whether ACCEL is less than 0, which means that the motor&#39;s rotation speed is decreasing e.g. because of a braking operation. In that case ACCEL is set to 0 in S 644 . 
   If ACCEL≧0 in S 642 , the routine then goes to S 646 , where the value of ACCEL is doubled. An ACCEL greater than 0 means that rotor  108  is being accelerated, for example as the motor comes up to speed. ACCEL is therefore predictively doubled because when a motor is started up, the rotation speed increases in accordance with an e-function, and if the doubling were not applied, the value of ACCEL would consequently be too low already after completion of the calculations. 
   Subsequent to S 644  and S 646 , the routine goes to S 648  where the value A* (equal to 400 μs, for example, because a period of approximately 400 μs is required for the commutation procedure) is added to the value of ACCEL (from S 644  or S 646 ). This value of ACCEL is then used in the RGL routine to modify the value of BW. The S 408  routine then ends at step S 652 . 
     FIG. 30  shows RGL routine S 410  for rotation speed control. This is based on a comparison between Hall time t_HALL and target time t_s, the latter corresponding to the desired rotation speed and being specified at input RA 0  of μC  40 . The controller according to the exemplary embodiment therefore does not work directly with rotation speed, but rather with the times needed by rotor  108  for a specific rotation angle. Hall time t_HALL corresponds to the time taken by the rotor to rotate 180° el. If rotor  108  has four poles and rotates at 3000 rpm, then
   t _HALL=60/(3000×4)=0.005  s= 5 ms  (6) 
Similarly, the time at 1000 rpm is
   t _HALL=60/(1000×4)=0.15  s= 15 ms  (7). 
   At low rotation speeds, actual value t_HALL is thus very large, for example 150 ms=0.15 s at 100 rpm, and is then substantially greater than target value t_s, which e.g. equals 5 ms at 3000 rpm. For this reason, system deviation RGL_DIFF in step S 654  is calculated as the difference (t_HALL−t_s) so that a positive result is obtained for the difference. 
   S 656  checks whether the system deviation is greater than a permitted positive maximum value RGL_DIFF_MAX. If so, then in S 658  the system deviation is set to that positive maximum value. This is important especially at start-up, when the system deviation would otherwise become very large. 
   If the response in S 656  is No, the program then goes to step S 660  and checks there whether the system deviation is less than a permitted negative maximum value −RGL_DIFF_MAX. If Yes, in S 662  the system deviation is set to that negative maximum value. (This refers to the situation in which the motor is faster than the desired rotation speed.) 
   Steps S 658 , S 660 , or S 662  are followed by S 664 , in which the calculation steps of a PI controller are performed. This involves multiplying the system deviation by a proportional factor RGL_P that can equal, for example, 2; the result is the proportional component RGL_PROP. 
   The system deviation is likewise multiplied by an integral factor RGL_I (equal, for example, to 0.0625), and is then added to the old integral component RGL_INT to yield a new integral component. 
   Lastly, length BW of a current block  444  or  446  ( FIG. 15 ) is calculated as the sum of the new proportional component and new integral component. 
   Proportional factor RGL_P and integral factor RGL_I are defined empirically as a function of the size of the motor and the inertia of the load being driven. 
   Since BW must not be longer than time t_HALL required by the rotor to rotate 180° el., the next step S 666  checks whether BW is too large; if so, in step S 668  the block length is limited e.g. Toe the instantaneous value t_HALL. 
   If the response in S 666  is No, routine S 410  goes to step S 670 , which checks whether BW is less than 0, meaning that the motor is running too fast. If so, in S 671  the value of BW is set to 0, i.e. no current flows to the motor. At the same time, integral component RGL_INT is set back to 0 (or to a low value). It has been found that this operation of setting the integral component back to a low value substantially improves the properties of the controller, especially with regard to overshooting of the set speed. 
   If the response in S 670  is No, in S 672  the block length is shortened to (BW−ACCEL), the value ACCEL being taken from S 648  of FIG.  29 . This value contains an acceleration-dependent component and the value A* (e.g. 400 μs) which was explained in FIG.  29 . The reason for S 672  is that during acceleration, e.g. at start-up, the next Hall change occurs earlier than at constant rotation speed, so that block length BW must be correspondingly shortened during acceleration. The doubling of value ACCEL in S 646  ( FIG. 29 ) also serves to make sufficient time available during acceleration for the commutation procedure, since as a motor starts up its speed increases approximately in accordance with an e-function, and this is taken into account in S 646 . 
   Using block length BW from S 672 , in S 673  times t_BLOCK_START and t_BLOCK_END, which are plotted in  FIG. 15 , are now calculated. In  FIG. 15 , t_BLOCK_START is the time span between t 1  and t 3 , and its magnitude is obtained from equation (3). Time t_BLOCK_END is obtained, as shown in  FIG. 15   d , by adding the value of BW to t_BLOCK_START. Times t_BLOCK_START and t_BLOCK_END are needed subsequently for the calculations in  FIG. 25  (COMMUT_CTRL routine), as has been explained in detail with reference to FIG.  26 . 
   If an “ignition advance” is desired, as has been explained with reference to equations (3a) and (4a), the formula
 
 t _BLOCK_START:= t _HALL+( t _HALL− BW )/2− VZ   (8)
 
is used in S 673 . VZ in this case is a constant equal to e.g. 400 μs, and its effect, as shown in  FIG. 15   d , is to shift the beginning of block  446  to t 3 ′, i.e. the current is switched on and shut off earlier; t 3 ′ can then be located before t 2 . The invention makes this possible because the reference point used for calculating t_BLOCK_START for transistors HSL  114  and LSR  136  is leading edge  370  of signal HALL, i.e. the edge before the previous one (cf. arrows  445  and  447  of FIG.  15 ).
 
   After S 673 , routine S 410  ends at S 674 . 
   The routine of  FIG. 30  thus yields a block length BW which becomes increasingly short as the actual rotation speed approaches the desired value. 
   The control function for block length BW interacts with the adaptive controller, described below with reference to FIG.  31  and  FIG. 32 , that further optimizes the value of BW by way of pulse duty factor pwm. BW should not exceed 95% of t_HALL so that time is available for the commutation procedure, and this is achieved by correspondingly modifying the PWM pulses of which a current block  444  or  446  ( FIG. 15 ) is composed; in other words, the average current in a block is raised or lowered by means of the adaptive controller. If BW is too long, the average current is automatically increased, by raising the pulse duty factor of these pulses, until block length BW has “shrunk” to a value which allows optimum execution of the commutation procedure. 
     FIG. 31  shows a MOD_pwm routine S 412  for modifying pulse duty factor pwm as a function of the motor&#39;s operating conditions. 
   Step S 900  checks whether block length BW generated in S 672  by the controller ( FIG. 30 ) is less than or equal to 50% of the instantaneous Hall time t_HALL. This (rotation-speed-dependent) value of 0.5*t_HALL represents a lower limit value below which BW should not substantially decrease in order minimize motor noise. The reason is that short drive current pulses cause the motor to produces more solid-borne sound, which is undesirable, and they also reduce efficiency. 
   If the value has fallen below the lower limit, S 902  checks whether pulse duty factor pwm is at least 10%. (It should not fall substantially below this value.) 
   If pwm is less than or equal to 10%, the program goes to step S 904  where pulse duty factor pwm_OUT at output RC 2  of μC  40  is set to the instantaneous value pwm; and then to S 906 , i.e. to the end of MOD_pwm routine S 412 . In this instance it is not possible to reduce pwm any further. 
   If pwm is greater than 10%, the program goes to step S 908 , which checks whether a counter PWM_CNT has a value of 0. This counter counts the number of times BW has reached or fallen below the lower limit value, i.e. 0.5*t_HALL, and at every fifth count value it causes pulse duty factor pwm to be reduced. 
   To achieve this, the μC has an internal 8-bit register which therefore has values between 1 and 256, and these values define pulse duty factor pwm of signal PWM outputted by μC  40  at its output RC 2 , which in this μC has a constant frequency of 20 kHz. Reducing the value in this internal register reduces pwm, and increasing the value in this register increases it. 
   If counter PWM_CNT has a value of 0 in S 908 , the program goes to step S 910  where this counter is set to a value of 5. Pulse duty factor pwm is then lowered in S 912  (cf. FIG.  22 ), thereby decreasing the mean value of motor current i 1 , i 2 . The program then goes to S 904 . 
   If counter PWM_CNT is not equal to 0 in S 908 , the program goes to step S 914 , where PWM_CNT is decremented by 1, i.e. in this case pwm does not change. 
   If the response in S 900  is No, the program goes to step S 916 , which checks whether block length BW calculated by controller RGL is too long, i.e. greater than or equal to 95% of t_HALL. This is undesirable because the commutation procedure requires approx. 400 μs, which would no longer be available if BW were too long. 
   If BW is not too long, the program goes to step S 904  (already explained), and pwm_OUT remains unchanged. 
   If BW is too long, the program goes to step S 918 , which checks whether pwm has already reached 100%; if so, the program goes directly to S 904 , since an increase above 100% is not possible, i.e. a continuous current then flows for the duration of BW. 
   If it is found in S 918  that the pulse duty factor is less than 100%, the next step is S 920 , where counter PWM_CNT is checked to determine whether its value is 0. If Yes, in S 922  counter PWM_CNT is set to 5. Value pwm is then incremented in S 924  (cf.  FIG. 21 ) so that the mean value of motor current i 1  or i 2  correspondingly increases. 
   If the response in S 920  is No, the program goes to step S 926 , where PWM_CNT is decremented by a value of 1; the routine then goes to step S 904 . 
     FIG. 32  explains the events in the flow chart of FIG.  31 . In  FIG. 32 , the abscissa shows relative block length b. This is defined as
   b=BW/t _HALL  (9) 
It therefore corresponds to the instantaneous ratio between block length BW and Hall time t_HALL, as a percentage. The ordinate shows the instantaneous pulse duty factor pwm, also as a percentage. As a reminder: t_HALL is the time required for rotor  108  to rotate 180° el. at the instantaneous rotation speed (cf. equations (6) and (7)).
 
a) Relative Block Length b Becomes Too High
 
   Let it be assumed that motor  100  is operating at an operating point C, namely at a block length BW equal to 80% of t_HALL, i.e. at b=80%, and at a pulse duty factor pwm of 35%. 
   When a load is placed on the motor, b increases along a characteristic curve  930  due to the action of controller RGL; pwm remains unchanged at 35%. At  932  the upper limit value b=95% is exceeded, and at  934  pulse duty factor pwm is increased by means of S 924  (FIG.  31 ), so that a higher average current flows, more energy is delivered to motor  100 , and its rotation speed rises. 
   The relative block length b is therefore reduced by rotation speed controller RGL at  936  and returns to the permissible range, but now with an increased pwm. (The increase in pwm is depicted in exaggerated fashion in  FIG. 32 ; it is performed only in small steps.) 
   Counter PWM_CNT prevents every minor excursion above upper limit value  932  from causing an increase in pwm. It has been determined empirically that an increase every fifth time yields very stable motor operation, but this factor can depend, for example, on the motor size, the type of load, etc. If this factor is too small, the controller tends to oscillate. Based on present understanding, values between 3 and 7 appear to be optimal. 
   b) Relative Block Length b Becomes Too Low 
     FIG. 32  shows, as a second example, an operating point D with a relative block length b=55% and a pwm of 80%. 
   As load on the motor is relieved, the characteristic curve follows a straight line  940  that falls below the lower limit value  942  (b=50%) and, at  944 , results in a relative block length b of approximately 47%. This causes an increase in motor noise, and is unfavorable in terms of motor efficiency. 
   Pulse duty factor pwm is therefore, by means of S 912 , reduced along a vertical line  946  (FIG.  32 ), thereby decreasing the mean value of the current delivered to the motor so that the rotation speed drops. 
   Rotation speed controller RGL ( FIG. 30 ) therefore calculates a greater block length BW so that relative block length b moves back, along a line  948 , into a range above lower limit value  942 . 
   When pronounced load changes occur, the operations just described can repeat several times. In principle, the rotation speed controller can adjust relative block length b and pulse duty factor pwm within the entire range enclosed by a dashed line  950  in  FIG. 32 , i.e. in this example a pwm between 10 and 100% and a relative block length b between 50 and 95%. This could also be referred to as an adaptive controller that always returns to the range defining its optimum efficiency and low motor noise. 
   Many variants and modifications are, of course, possible within the context of the present invention.