Abstract:
An apparatus displays pictures from sources having a plurality of horizontal frequencies. A scanning generator is operable at the plurality of frequencies and comprises an oscillator generating a signal. A divider with two selectable counts is coupled to the oscillator and divides the signal by a first count to generate a horizontal drive signal. A horizontal scanning amplifier generates a scanning signal responsive to the horizontal drive signal coupled thereto. A controller is coupled to the scanning amplifier and to the divider. In response to selecting another of the plurality of frequencies, the controller monitors the scanning signal and responsive to its presence inhibits selection of a second of the selectable counts. In the absence of the scanning signal the controller enables selection of the second of selectable counts and the divider generates a horizontal drive signal representative of the another one of the plurality of horizontal scanning frequencies.

Description:
This invention relates generally to the field of horizontal scanning systems for video apparatus and in particular to the control of systems operable at multiple horizontal scanning frequencies. 
     BACKGROUND 
     In a video display apparatus, scanning circuits are synchronized to a synchronizing component or sync derived from the input video signal. Hence, a video display apparatus which is operable at multiple horizontal scanning frequencies must be capable of synchronizing to a standard definition NTSC signal horizontal scanning frequency of nominally 15.734 kHz or to a high definition, Advanced Television Standards Committee, ATSC, signal having horizontal scanning frequency of nominally 33,670 kHz with 1080 active lines and interlaced scanning ( 1080 I). In addition to synchronizing to broadcast video signals, the apparatus may be required to display computer generated non-broadcast video signals, such as, for example, a super video graphics adapter signal or SVGA, having a horizontal frequency of 37,880 kHz. 
     Horizontal frequency oscillators employing phase locked loop control are widely known and used in video display apparatus. Dual and triple phase locked loops are also known and used to provide functional separation between potentially conflicting requirements of synchronization and scanning waveform generation. In a dual loop configuration, a first loop may be a conventional phase locked loop in which a voltage controlled oscillator output, or an output divided therefrom is compared with horizontal synchronizing pulses derived from the video signal to be displayed. The second phase locked loop, which for example, operates at the same frequency, compares the oscillator output from the first loop with a horizontal rate pulse, for example, a retrace pulse voltage derived from or representative of defection current flow. The error voltage from the second phase comparison is used to generate a width modulated pulse signal which determines the initiation of the deflection output device turn off, and subsequently, retrace initiation, or the phase of each line within the period of a vertical scan. 
     The response of the first phase locked loop may be optimized for fringe area reception of broadcast video signals suffering poor signal to noise ratios. Such signals suggest that the response of the first phase locked loop is relatively slow. Accordingly, the first loop may have a narrow bandwidth to optimize phase jitter reduction. However, since a video display apparatus required to be operable with signals from a variety sources and with differing horizontal frequencies. The response of the first phase locked loop represents a compromise between a narrow bandwidth for minimized phase jitter and a wide bandwidth, fast loop response capable of rapid phase recovery. For example, a narrow bandwidth loop is suited to synchronization by low noise, non-broadcast computer generated signals, whereas and wide bandwidth, fast loop response, capable of rapid phase recovery is required for synchronization of video cassette recorder (VCR) replay signals where abrupt changes in horizontal sync. pulse phase, by as much as 10 microseconds may occurring between the beginning and end of the vertical banking interval. Hence tradeoffs in respective loop responses may be made to provide adequate weak signal performance without significant overall degradation of receiver performance. The second phase locked loop generally has a faster loop response. Accordingly, the second phase locked loop may have a wider bandwidth allowing it to track variations in the defection current due to horizontal output transistor storage time variations, or high voltage transformer tuning effects. Such tight tracking yields a straight, non-bending raster largely independent of beam current loading. 
     The use of voltage controlled oscillators for horizontal frequency signal generation is well known. It is known to employ an oscillator operating at a multiple of the input horizontal sync. frequency and to achieve synchronization by means of counters with a selectable counts. However, immediate scanning circuitry failure results when scanning frequency current is interrupted by count selection while scanning. 
     SUMMARY OF THE INVENTION 
     Scanning circuitry failure resulting from count selection while scanning is advantageously prevented by an inventive arrangement. A scanning generator operable at a plurality of horizontal scanning frequencies comprises an oscillator generating a signal. A divider with at least two selectable counts is coupled to the oscillator and divides the signal by a first count to generate a horizontal drive signal. A horizontal scanning amplifier generates a scanning signal responsive to the horizontal drive signal coupled thereto. A controller is coupled to the scanning amplifier and to the divider. In response to selection of another one of the plurality of horizontal scanning frequencies, the controller monitors the scanning signal and responsive to a presence the controller inhibits selecting a second of the least two selectable counts. In the absence of the scanning signal the controller enables selection of the second of the least two selectable counts and the divider generates a horizontal drive signal representative of the another one of the plurality of horizontal scanning frequencies. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a block diagram of an exemplary horizontal frequency oscillator employing three phase locked loops with various inventive arrangements. 
     FIG. 2 is a schematic diagram of part of FIG.  1  and shows an inventive switched active filter. 
     FIG. 3 shows a voltage controlled oscillator including inventive features which form part of FIG.  1 . 
     FIG. 4 is a schematic diagram of the inventive switching interlock which forms of part of FIG.  1 . 
     FIG. 5A is a plot illustrating the gain versus frequency characteristic of the inventive switched active filter of FIG.  2 . 
     FIG. 5B is a plot illustrating the phase versus frequency characteristic of the inventive switched active filter of FIG.  2 . 
    
    
     DETAILED DESCRIPTION 
     A horizontal frequency oscillator and deflection amplifier employing three phase locked loops and operable at a plurality of frequencies is shown in FIG.  1 . In a first phase locked loop  10 , an input video display signal, for example a standard definition NTSC signal is coupled to a sync separator, SS, where a horizontal synchronizing signal component is separated. A voltage controlled oscillator has a frequency of 32 times an NTSC horizontal frequency, 1 Fh, and is divided by 32 in a counter, depicted as, ÷32. The divided oscillator signal is coupled as one input to a phase detector PD, with the second input coupled to the separated sync component. The resulting phase error between the divided oscillator signal and the separated sync component is coupled from phase detector, PD, to synchronize the 32Fh voltage controlled oscillator. The functional elements of PLL  10 , form part of a bus controlled integrated circuit, for example type TA 1276 . The standard definition horizontal sync component from PLL 10  is coupled to a sync source selector switch SW 15  which provides selection between a plurality of synchronizing signals coupled as input sources to synchronize second and third controlled horizontal oscillator loops,  100  and  410  respectively. Selector switch SW 15  is depicted with three exemplary sync sources, namely a standard definition NTSC sync signal, a high definition sync signal, for example ATSC  1080 I, and a computer generated SVGA sync signal, however, sync selection for horizontal oscillator synchronization signal is not limited to these examples. Sync switch SW 15  is controlled by switching signal  15   a  which is generated by microcontroller  800  in response to a user control command, for example, as generated by a remote control transmitter RC, which communicates by wireless means to receiver IRR,  801  which input the remote control data to microcontroller  800 . Remote control RC allows display signal source selection, for example, changing broadcast TV channels between HD and SD broadcasts or viewing a computer program with selectable display resolution. 
     The three phase locked oscillators depicted in FIG. 1 are advantageously controlled to provide optimized performance, to only with input signals of differing frequencies but also with signals subject to timing perturbations. During the display of NTSC signals, loops  10 ,  100  and  410  are utilized. However NTSC signals may originate from a broadcast source or a VCR. The latter source may be subject to sync phase perturbations, thus such signal disturbances are advantageous accommodated within PLL  100  by means of controlled selection of low pass filter characteristic. Selection of high definition signal inputs, for example ATSC or SVGA cause PLL 10  to be bypassed reducing the sync system to two controlled loops, for example PLL 100  and PLL 410 . Thus microcontroller  800  is required to control input video display selection responsive to user commands, to control sync source selection responsive to the display selection, control the oscillator frequency, the oscillator divider and phase locked oscillator low pass filter characteristics. 
     The selected synchronizing signal  5 , from switch  15 , is coupled to an input of phase detector  50  to facilitate synchronization of the second phase locked loop  100 . A second input to phase detector  50  is supplied with signal  401 , derived by division of voltage controlled oscillator signal  301 . The resulting phase error signal  11  is low pass filtered and applied to control VCO  300  thus achieving synchronism with the input video display signal horizontal sync. The third phase locked loop  410  compares a signal from voltage controlled oscillator VCO  300  with a scanning related signal Hrt, for example a horizontal scan derived pulse resulting from a scanning current generated by a scanning amplifier  500 . 
     The center frequency of horizontal oscillator  300  is determined by means of control bus  420 , for example an I 2 C bus, which advantageously changes the oscillator frequency and the low pass filter characteristics. In addition an advantageous protection circuit  600  prevents circuitry damage resulting from accidental, erroneous and undesired divider switching during scanning by means of an electronic interlock. 
     Operation of the second and third horizontal oscillator loops and scanning amplifier of FIG. 1 is as follows. A horizontal sync signal  5 , depicted as an exemplary positive pulse, is selected by switch  15  from either PLL 10  or sync signals derived from a plurality of input display signals. Synchronizing signal  5  is applied to a phase detector  50  where it is compared with a horizontal rate signal  401  produced by division of line locked clock signal LLC,  301  from voltage controlled oscillator, VCO  300 . Block  400  represents an exemplary deflection processing integrated circuit IC  400 , for example type TDA 9151 . Integrated circuit  400  is bus controlled, for example by I 2 C bus  420 , and also includes a phase detector PLL 3 , and dividers  415  and  415 A. Divider  415 A is controlled by signal  402 , to provide division ratios of  432  and  864  respectively and thereby produce horizontal rate signals in two bands of frequencies, nominally 1 Fh and 2 Fh. Control signal  402  is coupled to switch  412  which inserts or bypasses divider  415 A, to provide two division ratios. Thus voltage controlled oscillator, VCO  300  operates only in a band of frequencies about 13.6 MHz, but is synchronized to horizontal frequencies differing by more than 2:1. Examples of such non-integer related horizontal frequencies are NTSC signals where the horizontal frequency, represented by 1 Fh, is 15,734 kHz and an ATSC  1080 I signal with a horizontal frequency, represented relative to the NTSC signal as 2.14 Fh, or 33,670 kHz. During the display of NTSC derived images, switch  412  selects divider  415 A which provides a division ratio of 864:1 yielding a frequency nominally that of the NTSC horizontal frequency 1 Fh. Similarly for the display of images with horizontal frequencies of 2 Fh or greater, for example an ATSC  1080 I signal, switch  412  bypasses divider  415 A resulting in a division ratio of  432  which produces a horizontal frequency 2 Fh, of 31,468 kHz, twice that of the NTSC standard. However, the ATSC  1080 I horizontal frequency is not an integer multiple of the NTSC signal 1 Fh and is actually 2.14 times the NTSC frequency. Thus to achieve synchronism with a  1080 I input signal, or any non 2 Fh sync rate, requires that the VCO frequency is changed to a frequency which when divided by  432  yields a frequency which may be synchronized with that of ATSC  1080 I, or the selected input signal horizontal rate. 
     Divided line locked clock signal  401  is also coupled to synchronize the third loop  410  by means of phase detector PLL 3 , which compares clock signal  401  with a scan current derived pulse Hrt,  501 . An output signal  403  from PLL 3  is coupled via a driver stage  450  to a horizontal scanning stage  500  which generates a scan related current, for example, in a display device or an electron beam deflection coil. In addition to coupling to PLL 3 , scanning pulse Hrt is also coupled to protection circuit  600  and X-ray protection circuit  690 . 
     As has been described, the operating frequency of the second and third phase locked loops may be changed in the ratio of 2:1 by means of divider switching. However, to achieve synchronization of the VCO at other than harmonically related frequencies, for example with an ATSC  1080 I frequency of 2.14 F, or an SVGA signal with an 2.4 Fh horizontal frequency, requires that the second phase locked loop controls the VCO to achieve a non-integer horizontal frequency nominally of between 2.14 and 2.4 times that of an NTSC horizontal frequency. In voltage controlled oscillator  300  an advantageous frequency setting DC potential, FREQ. SET,  302  determines an oscillator frequency which when divided generates a nominal horizontal frequency. The frequency setting DC potential is generated by a digital to analog converter and is applied to a voltage variable capacitor or varicap diode which forms part of the oscillator frequency determining network. The oscillator is synchronized to the input sync signal by means of a phase detector error signal, which is filtered and applied to an inductor which is part of the frequency determining network of VCO  300 . In simple terms, a frequency setting DC is applied to the varicap diode end of the series tuned network, with the phase error signal applied at the inductor end. Thus frequency and phase control signals are applied across the frequency determining tuned circuit. 
     Voltage controlled oscillator  300  is depicted in FIG.  1  and is shown in schematic form in FIG.  3 . Operation of the advantageously controlled oscillator  300  is as follows. Microcontroller  800  and a memory, (not shown), access and output frequency setting data via data bus  420 , for example an I 2 C bus, as illustrated in FIG.  1 . The I 2 C bus is connected to a digital sync processor  400 , to provide various control functions, and to a digital to analog converter  700  which separates and converts data into analog voltages. Digital to analog converter  700  generates frequency switching control signal  1 H_SW,  701 , and VCO frequency setting voltage FREQ. SET  302 . In FIG. 3, the frequency setting voltage FREQ. SET  302  is coupled via a resistor R 1  to the junction of resistors R 3 , R 4  and a capacitor C 3 , which in conjunction with resistor R 1  forms a low pass filter to ground. Resistors R 1  and R 3  form a potential divider for the frequency setting voltage with resistor R 3  connected to DAC  700  reference voltage (Vref). Thus analog voltage  302  is nominally halved and referenced to the DAC reference voltage (Vref) to apply a nominal voltage of about +3.8 volts of biasing potential to varicap diode D 1 . The junction of resistors R 1 , R 3  and capacitor C 3  are coupled to the cathode of varicap diode D 1  via a resistor R 4 . Thus the nominal DC voltage value, derived from voltage (Vref), plus a data determined frequency setting voltage  302 , from ADC  700 , are applied to the varicap diode D 1  of the oscillator frequency determining network. The frequency setting voltage  302 , is nominally zero volts in 1 Fh and 2 Fh modes and rises to about +7 volts when operation at 2.4 Fh, for example SVGA, is selected. 
     The oscillator is formed by PNP transistor Q 3  which has the emitter connected to a positive supply via a resistor R 7  and the collector connected to ground via a parallel combination of a resistor R 8  and a capacitor C 4 . The base of transistor Q 3  is connected t the positive supply via a resistor R 6 , and is coupled to ground via a capacitor C 5 . The oscillator frequency is determined largely by a series resonant network formed by an adjustable inductor L 1  and a parallel combination of varicap diode D 1  and capacitor C 4 . The junction of resistor R 4 , diode D 1  cathode and capacitor C 4  are coupled to the base of transistor Q 3  via capacitor C 6 . The collector of transistor Q 3  is connected via capacitor C 8  to the junction of inductor L 1  and a resistor, depicted in FIG. 2 as R 6 , which supplies the processed phase error signal  201  for oscillator synchronization. Thus, the frequency control and the phase synchronization signals are applied across the series resonant network formed by elements D 1 , C 4 , L 1 . Initial tuning of the oscillator may be achieved by setting the DAC voltage  302  to nominally zero volts, and with a 1 Fh horizontal frequency sync signal coupled to the phase detector inductor L 1  is adjusted to center the phase detector error signal within its operating range. In an alternative oscillator setting method a fixed non-adjustable inductor L 1  is employed. A horizontal frequency sync signal of 1 Fh is applied to the phase detector and DAC voltage  302  is varied until the phase detector error signal is centered. The data value corresponding to this centering value of voltage  302  is then stored. To determine the frequency set voltage for operation at 2.4 Fh, the immediately preceding method is repeated with the data value which centered the loop being stored. The oscillator output signal is extracted from the emitter transistor Q 3  at resistor R 7  and coupled to the emitter of PNP transistor Q 4  via a coupling capacitor C 6 . Transistor Q 4  is configured as a grounded base amplifier with the base decoupled to ground by a capacitor C 7  and connected to a positive supply via a resistor R 11 . The collector of transistor Q 4  is connected to ground via resistor R 10 . Thus the oscillator output signal is developed across resistor R 10  and coupled to the sync processing IC  400  as a line locked clock, LLC  301 . 
     Frequency switching signal SEL. H. FREQ.,  202 / 402  which controls divider  415   a  selection is also coupled to inventive low pass active filter  200 , which is shown in FIG.  2  and functions follows. A phase error signal Φ ERROR,  11 , which results from the phase comparison between signal  401 , divided VCO, and input signal sync  5 , is coupled to input resistor R 1 . Input resistor R 1  is connected in series with resistor R 2  to a inverting input of an integrated circuit amplifier  210 . The junction of resistors R 1  and R 2  is connected to a fixed contact 1 Fh of switch S 1 . The moving contact of switch S 1  is connected to the junction of a parallel combination of resistor R 3  and capacitor C 3  and a parallel combination of resistor R 4 , and capicitor C 4 . Negative feedback is applied from the output of amplifier  210  to the inverting input via a frequency dependent network formed by capacitor C 2  and series connected combination of parallel networks of resistor R 4  and capacitor C 4  and resistor R 3  and capacitor C 3 . Parallel network R 3 , C 3  is connected between switch S 1  wiper and the inverting input of amplifier  210 . When switch S 1  selects position 1 Fh, resistor R 2  is connected in parallel with the parallel combination of resistor R 3  and capacitor C 3  with the result that the newly formed parallel network, R 2 , R 3 , C 3  has little effect in the determination of the amplifier gain or frequency response. Thus when synchronized at 1 Fh, with switch position 1 Fh selected the amplifier gain is set by input resistor R 1 , with the frequency response determined by capacitor C 2  and parallel network R 3 , C 3 . When the display is operating at a horizontal frequency greater than 1 Fh switch S 1  selects position 2 Fh and resistor R 2  becomes the predominant gain determining component, with the frequency response controlled by the series combination of capacitor C 2  and parallel networks R 3 , C 3  and R 4 , C 4 . The non-inverting input of amplifier  210  is biased to a positive potential of about 2.5 volts. 
     The output from amplifier  210  is coupled via series connected resistors R 5  and R 6  to form a processed phase error signal, PROC. Φ ERROR,  201 , for coupling to synchronize VCO  300 . The junction of resistors R 5  and R 6  is decoupled to ground by a capacitor C 1  which forms a low pass filter to prevent high frequency noise generated, for example by switched mode power supply operation from producing spurious VCO phase modulation. The junction of resistors R 5  and R 6  is also connected to a peak to peak limiter or clipper formed by the emitters of PNP transistor Q 1  and NPN transistor Q 2 . The collector of transistor Q 1  is connected to ground with collector of transistor Q 2  connected to a positive supply via a resistor R 9 . The base of transistor Q 2  is connected to the junction of series connected resistors R 10  and R 7 . Resistor R 10  is connected to ground and resistor R 7  is series connected to a further positive supply via a resistor R 8 . The junction resistors R 7  and R 8  is connected to the base of transistor Q 1 . Thus, resistors R 7 , R 8  and R 10  form a potential divider which determines the peak to peak clipping values of approximately +0.3 v and +2.2 volts at which processed error signal  201  is limited. 
     In a phase locked loop, the selection of phase detector output filtering is a compromise between static and dynamic performance. For example, synchronization to a computer generated SVGA signal may require, or may benefit from, a narrow bandwidth VCO control signal, which will provide a highly phase stable oscillator and horizontal frequency. However, as described previously, VCR replay sync signals may include abrupt horizontal sync phase changes in the vicinity of the vertical sync and vertical blanking intervals. To prevent, or mitigate, the effect of this phase change requires that the loop have a wider bandwidth than required for either computer generated SVGA signals or broadcast signals which are not subject to abrupt phase disturbances. Advantageous amplifier  210  is arranged as an active low pass filter where output signal components are feedback to the inverting input via frequency dependent series connected network C 2 , C 3 , C 4 , and R 3 , R 4 . Advantageously switch S 1  is controlled responsive to a selected horizontal oscillator frequency such that in switch position 1 Fh, resistor R 2  is connected in parallel with parallel combination R 3 , C 3  to form an impedance in series with the inverting input. This parallel combination of resistors R 2 , R 3  and C 3  produces little effect on filter gain or frequency response. In switch position 1 Fh the filter gain is determined by the impedance of network C 2 , C 1  and R 4  divided by the value of input resistor R 1 . Clearly as the loop operating frequency approaches DC the impedance of capacitor C 2  becomes large and the loop gain approaches an upper limit condition as depicted in FIG.  5 A. When operating at other than 1 Fh horizontal frequency switch S 1  is controlled to select position 2 Fh. In switch position 2 Fh filter gain is determined by the impedance of feedback network R 3 , C 2 , C 1  and R 4 , divided by the series combination of resistors R 1  and R 2 . Since resistor R 2  is significantly larger than resistors R 3  the gain in the 2 Fh is reduced relative to that of switch position 1 Fh. Thus the active filter gain and bandwidth are controlled to be different in response to a selection of horizontal operating frequency. 
     During operation at a horizontal frequency of 2 Fh or higher, switch S 1  selects the 2 Fh position with the result that the gain at frequencies close to DC is approximately 10 dB, as is illustrated by the broken line in the amplitude versus frequency plot of FIG.  5 A. The gain then falls to zero at about 10 Hz and continues to fall reaching −20 dB at about 100 Hz. Thus when operating in a 2 Fh mode with switch S 1  in the 2 Fh position the zero gain bandwidth is approximately 10 Hz. FIG. 5B shows phase versus frequency plots for the two horizontal frequencies with the 2 Fh mode indicated by a broken line. When operating at an NTSC frequency of 1 Fh, switch S 1  is controlled to select the 1 Fh position which increases the filter gain and provides a zero gain bandwidth in excess of 10 kHz. Reference to FIG. 5A illustrates that greater low frequency filter gain is employed during operation at 1 Fh than that used during operation at higher horizontal frequencies. In addition the filter produces a significantly wider phase error signal bandwidth than that obtained in the 2 Fh mode. Active filter gain and frequency response switching is advantageously achieved with a single switch contact which provides savings in printed circuit board area which consequently reduces susceptibility stray field pickup and spurious phase instability. 
     User input signal selection results in a corresponding selection between the plurality of horizontal frequencies which is communicated by microcontroller  800  to sync source selector switch  15  and sync processing IC  400 . Microcontroller  800  generates a specific control command LFSS, which is addressed to sync processing IC  400  to start or stop horizontal and frame generation. Thus horizontal drive output signal,  403 , may be terminated by processor  800  control as depicted by output switch  412   a . Hence, in the absence of horizontal drive signal  403 , horizontal scan amplifier  500  ceases to generate scanning current and consequently pulse Hrt is no longer produced. Following the horizontal off command (LFSS), the microcontroller transmits control words addressed to the digital to analog converter DAC  700 . A first DAC  700  control word may represent a horizontal frequency switch command which is output from DAC  700  as analog control signal  1 H_SW,  701 , and coupled to switching interlock  650 . The DAC may also receive a second control word which generates an analog frequency setting potential FREQ. SET  302 . 
     The microprocessor generated command LFSS which turns off horizontal drive  403 , consequently terminates generation of pulse Hrt. The absence of pulse Hrt indicates the cessation of scanning which allows control signal  1 H_SW to be coupled to form frequency switching signal SEL. H. FREQ. Thus signal SEL. H. FREQ.  402 , is able to change state thereby selecting a different division ratio within sync processor  400  and hence a different horizontal frequency for loops  100  and  410 . Since scanning is terminated by command LFSS, divider  415 A may be inserted or bypassed from the divider chain, without causing damage to the horizontal driver  450  or horizontal scan amplifier  500 . The microcontroller transmits the horizontal off command, prior to transmitting a horizontal frequency switch command to ensure that horizontal scanning amplifier  500  is quiescent and thereby avoid circuitry damage. However, an advantageous protection circuit  600  provides a further level of protection by monitoring to determine that digital command LFSS, generated by the microprocessor and transmitted by I 2 C bus was demultiplexed and implemented by sync processor  400 . Thus protection circuit  600  verifies implementation of the bus instruction and allows horizontal frequency selection to occur in the absence of horizontal scan pulses Hrt. In addition sync processor  400  and scanning amplifier  500  are protected against erroneous divider changes resulting from spurious signals generated, for example by, DAC  700 , errant circuit functions, power supply loading or CRT arcing. 
     Advantageous protection circuit  600 , is shown in FIG. 4, which provides various control functions related to the presence or absence of scanning current as indicated by the detection of pulse Hrt,  501 . Circuit block  610 , detects the presence or absence of pulse  501  and generates an active low interrupt, SCAN-LOSS INTR.  615 , which is coupled to microcontroller, μ CONT.  800 . 
     In a second protective function, circuit  600  verifies that sync processor instruction LFSS has terminated horizontal drive generation as indicated by the absence of pulse Hrt. Thus by interlocking horizontal frequency selection with scanning presence, frequency switching is inhibited in the presence of pulse Hrt. Horizontal frequency selection data is coupled from microcontroller  800  by bus  420 . The bus is demultiplexed and the frequency selection data is digital to analog converted by DAC  700  to form switching signal  1 H_SW for coupling to circuit block  650 . The circuitry of block  650  allows the logical state of signal  1 H_SW to be coupled for frequency selection only if scan amplifier  500  is not generating pulses Hrt. Thus horizontal frequency change is interlocked and prevented until scan related pulses cease. 
     In block  610  of FIG. 4, scan derived pulses Hrt are rectified by diode D 1  and charge capacitor C 1  positively via a resistor R 2  towards the positive supply. The junction of resistor R 2  and capacitor C 1  are joined to the base of a PNP transistor Q 1  with the result that the positive charge developed across capacitor C 1  turns the transistor off when deflection related pulses are present. The emitter of transistor Q 1  is coupled to a positive voltage supply via a diode D 2  which prevents base emitter zenner breakdown and ensures that transistor Q 1  turns off when the pulse derived charge across capacitor C 1  is approximately 1.4 volts or less. The collector of transistor Q 1  is coupled to ground via resistors R 3  and R 4  connected in series. The junction of the resistors is coupled to the base of an NPN transistor Q 2  which has the emitter grounded and the collector coupled via a resistor R 7  to form an open collector output signal. Thus when pulses Hrt are present transistor Q 1  is turned off, which in turn turns off transistor Q 2  rendering output signal  615 , scan loss interrupt, an open circuit. When scan related pulses are absent, as a consequence for example, bus derived control function, circuit failure or X-ray protection, the positive charge developed across capacitor C 1  is dissipated via the series combination of resistors R 1  and R 2  thus allowing capacitor C 1  to charge towards ground potential. When the potential across capacitor C 1  is nominally 1.4 volts transistor Q 1  turns on with the collector terminal assuming the nominal potential at the cathode of diode D 2 . Thus this positive potential of about 7 volts at transistor Q 1  collector is applied via the potential divider formed by resistors R 3  and R 4  to the base of transistor Q 2 , which turns on taking the collector and output signal  615  to nominal ground potential. Signal  615  is an interrupt signal which, when low, signals microcontroller  800  that scanning current is absent in the exemplary display or coil. 
     The collector of transistor Q 1  of FIG. 4, is also coupled to circuit block  650  which advantageously allows or inhibits changes of horizontal frequency originated by the microcontroller and communicated as a data word via bus  420  to a digital to analog converter DAC  700 . The digital to analog converter  700  converts the data word and generates an analog control signal  1 H_SW which has two exemplary voltage values. When control signal  1 H_SW is nominally at zero volts (Vcesat), divide by two stage of processor  400  is bypassed and divider  415  divides the VCO output signal LLC,  301 , by  432  to produce a frequency in a higher band of horizontal frequencies equal to or greater than 2 Fh. When control signal  1 H_SW is approximately 9.6 volts, divide by two stage  415 A is selected which produces a combined division of  864 . Thus the VCO generated line locked clock LLC  301  is divided by  864  to produce a nominal frequency of 1 Fh. The collector of transistor Q 1  is coupled via series connected resistors R 5  and R 6  which form a potential divider to ground. The junction of resistors R 5  and R 6  is coupled to the base of an NPN transistor Q 3  which has a grounded emitter. The collector of transistor Q 3  is connected to the positive supply via a load resistor R 8  and is also coupled to the base of an NPN transistor Q 4  via a resistor R 10 . The emitter of transistor Q 4  is coupled to the junction of a potential divider formed between the positive supply and ground where resistor R 9  is connected to the supply and resistor R 11  is connected to ground. Thus, the emitter of transistor Q 4  is biased at about 4 volts. Hence transistor Q 4  is turned on when the base voltage exceeds about 4.7 volts causing the collector to assume the nominal emitter potential. The collector of transistor Q 4  is connected directly to the junction of control signal  1 H_SW, and both the trigger input TR and threshold the threshold input of input TH of integrated circuit U 1 , for example I.C. type LMC  555 . Thus with both the trigger and threshold inputs clamped to 4 volts, changes in control signal  1 H_SW resulting from bus generated command or erroneous signal pickup are prevented from changing the output state of I.C. U 1 . The threshold input of integrated circuit U 1  responds when voltage value of control signal  1 H_SW exceeds about 5.3 volts and results in the selection of 1 Fh scanning frequency. The trigger input of I.C. U 1  responds to a negative transition of control signal  1 H_SW when the voltage value is less than approximately 2.6 volts which results in the selection of 2 Fh scanning frequency. 
     Operation of circuit  650  is as follows. The presence of Hrt pulses coupled to circuit  610  turns off transistor Q 1  with the collector assuming a nominally ground potential via the parallel combination of series connected resistors R 3  and R 4 , and series connected resistors R 5  and R 6 . Thus, transistor Q 3  is also turned off with the collector assuming the nominal supply voltage via resistor R 8 . This positive potential is applied to the base of transistor Q 4  which turns on connecting the junction of control signal  1 H_SW and integrated circuit U 1  to a potential of about +4 volts. With +4 volts applied to both the trigger and threshold inputs of IC U 1 , U 1  is prevented from responding to changes of control signal  1 H_SW. Thus the current status of select horizontal frequency control signal  202 / 402  is maintained and cannot be changed whilst scanning pulses Hrt are present. In the absence of scanning pulses transistor Q 1  turns on and the collector assumes the nominal supply potential. This positive potential is coupled via series resistors R 5  and R 6  and turns on transistor Q 3  which in turn, turns off transistor Q 4 . With transistor Q 4  off, the inhibit is removed from integrated circuit U 1  thus, for 1 Fh operation signal  1 H_SW assumes a high voltage value, and IC U 1  assumes a low voltage value. Similarly when 2 Fh operation is selected control signal  1 H_SW assumes a low voltage with U 1  output assuming a high voltage value. Thus, change of horizontal frequency is prevented when scan related pulses Hrt are present, thereby preventing probable failure of horizontal scanning stage  500 . 
     In circuit block  655  of FIGS. 1 and 4, integrated circuit U 1  advantageous provides a further protective function by controlling power supply selection to ensure that a higher voltage power supply is enabled only when horizontal scanning frequencies of 2 Fh or greater are selected. In addition circuit  655  prevents unwanted control instructions or spurious signals from enabling the higher voltage supply during scanning at standard definition rates. Such erroneous activation of the higher voltage canning supply causes probable destruction of scanning amplifier  500 . 
     In FIG. 4, a power supply switching command  2 H_VCC, from DAC  700 , is coupled to series connected resistors R 13  and R 14  which form a potential divider to ground. The junction of the resistors is connected to the base of a transistor Q 5  which has the emitter grounded and the collector connected as an open collector output to generate power supply control signal SEL.  1 H_VCC,  656 . The base of a transistor Q 5  is also connected to a discharge output of I.C. U 1 . The operation of circuit block  655  is as follows. In response to selection of scanning frequencies of 2 Fh or greater, a power supply enabling command is generated by microcontroller  800  and transmitted by bus  420 . The power supply enabling command is demultiplexed by digital to analog converter DAC  700  which generates a power supply control signal  2 H_VCC,  702 . When control signal  702  is high, for example, approximately +9.6 volts transistor Q 5  is turned on and the collector, and power supply control signal SEL.  1 H_VCC,  656  assumes a potential of nominally zero volts, (Vcesat) of transistor Q 5 . Thus with power supply control signal  656  low, a higher voltage supply is enabled for scanning operation at higher horizontal frequencies. However enablement of the higher voltage power supply is advantageously controlled or interlocked to prevent erroneous activation of the higher voltage power supply during scanning at NTSC rates. Such erroneous power supply enablement generates excessive scanning current, increases retrace pulse Hrt amplitude, and consequently causes failure of scanning amplifier  500 . 
     Transistor Q 5  is advantageously controlled by a discharge output circuit of IC U 1  which assumes a saturated, low impedance state during scanning operation at NTSC rates. Thus the discharge output circuit of IC U 1  prevents erroneous high voltage supply activation by clamping transistor Q 5  base to nominal ground potential during scanning at 1 Fh rates, inhibiting generation of signal  2 H_VCC. Thus enablement of the higher voltage power supply is prevented and signal SEL.  1 H_VCC,  656  remains high, sustaining a 1 Fh power supply condition having a lower operating voltage. The discharge circuitry of I.C. U 1  becomes inactive when the output circuitry of U 1  changes state, i.e. output signal SEL H. Freq. goes high in response to the selection of a 2 Fh operating mode. 
     The inventive interlock between horizontal frequency selection and scanning presence advantageously verifies implementation of bus transmitted instruction. In addition circuit damage is averted by the inventive circuit which prevents erroneous horizontal frequency switching or power supply activation.