Abstract:
A novel frequency scheme for a police radar detector enables improved sweeping of the X, K, K u  and K a  radar bands. The novel frequency scheme requires two initial frequency conversions for detection of the X, K and K a  radar bands and a single initial frequency conversion for the K u  radar band with single initial frequency conversion being enabled by disabling the second mixer. During sweeping of the X, K and K a  bands, selectable, i.e., upper or lower, sideband suppression is employed to reduce undesired image sidebands and noise prior to the second frequency conversion. In addition, noise at the second IF frequency is reduced to prevent this noise from feeding through the second mixer into the second IF amplifier. During the K u  band sweep, the second mixer is bypassed and shunting of signals at the second IF frequency is disabled so that these signals enter the second IF amplifier.

Description:
BACKGROUND OF THE INVENTION 
     The present invention relates in general to police radar detectors and, more particularly, to police radar detectors and methods of operating police radar detectors wherein an improved frequency scheme enables improved sweeping of the X, K, K u , and K a  radar bands, and also selective sideband suppression during sweeps of the X, K and K a  radar bands. 
     Police have used radar waves to monitor the speed of motor vehicles for many years. The frequencies currently used throughout the world include: the X band—10.50 gigahertz (Ghz) to 10.55 Ghz; the K u  band—13.40 to 13.50 Ghz; the K band—24.05 Ghz to 24.25 Ghz; and the K a  band—33.40 Ghz to 36.00 Ghz. To alert motorists of the presence of police radar, electromagnetic signals within these radar bands are monitored using a police radar detector which issues an audible and/or visual alert upon detection of a radar signal within one of the bands. 
     Police radar detectors are basically superheterodyne radio receivers in which the tuning of the receivers is repeatedly swept through the frequencies which are to be received or detected, i.e., the frequencies of the radar bands. A typical superheterodyne radio receiver includes an antenna for receiving electromagnetic signals and a circuit for mixing signals from the antenna and local oscillator (LO) signals to convert the frequency of received electromagnetic signals to the frequency of an intermediate frequency (IF) with the LO being swept in frequency to tune the required frequencies. Incoming electromagnetic signals can then be received at the LO frequency plus or minus the IF frequency, known as upper and lower sidebands, respectively. 
     Generally, signal reception occurs in only one of these two sidebands and the other sideband or image frequency is suppressed by filtering or phasing to thereby perform single sideband (SSB) reception. SSB operation is usually preferable because it generally delivers improved noise figure relative to double sideband (DSB) operation and also reduces sensitivity in the alternate sideband, thus reducing susceptibility to undesired signals. Sometimes DSB operation is intentionally adopted in the interest of economy or because in some circumstances it can facilitate expanded frequency coverage. 
     The frequencies used within the police radar detector, including frequencies or swept frequency bands of local oscillators and frequencies of intermediate amplifiers, and the tuning methods, are referred to in the art as frequency schemes and a variety of frequency schemes are known and utilized in police radar detectors. For example, see U.S. Pat. No. 5,068,663; 5,268,689; 5,305,007; and, 5,917,441. 
     While known frequency schemes are satisfactory for operation of police radar detectors, there is an ongoing need for new and advantageous frequency schemes which improve operation of police radar detectors, reduce costs of manufacturing police radar detectors and/or simplify circuitry or operation of police radar detectors. 
     SUMMARY OF THE INVENTION 
     This need is currently met by the invention of the present application wherein a novel frequency scheme for a police radar detector enables improved sweeping of the X, K, K u  and K a  radar bands. The novel frequency scheme requires two initial frequency conversions for detection of the X, K and K a  radar bands and a single initial frequency conversion for the K u  radar band with single initial frequency conversion being enabled by disabling the second mixer of the police radar detector. During sweeping of the X, K and K a  radar bands, selectable sideband suppression, i.e., either upper sideband suppression or lower sideband suppression, is employed to reduce undesired image sidebands and noise prior to the second frequency conversion. In addition, noise at the second intermediate frequency, for example about 725 megahertz (Mhz), is reduced to prevent this noise from feeding through the second mixer into the second IF amplifier. During the K u  radar band sweep, the second mixer is bypassed and shunting of signals at the second IF frequency is disabled so that these signals enter the second IF amplifier. 
     In accordance with one aspect of the present invention, a police radar detector comprises an antenna for receiving incoming electromagnetic signals. A first local oscillator generates a first local oscillator signal which is swept through a first range of frequencies to sweep the X, K and K a  radar bands and a second range of frequencies to sweep the K u  radar band. A first mixer is coupled to the antenna and the first local oscillator for mixing the incoming electromagnetic signals with the first local oscillator signal to generate first intermediate frequency signals. A second local oscillator generates a second local oscillator signal. A second mixer is coupled to the first mixer for mixing first intermediate frequency signals with the second local oscillator signal to generate second intermediate frequency signals at a second intermediate frequency. Detector circuitry is coupled to the second mixer for detecting received electromagnetic signals within the X, K u , K and K a  radar bands. Signal conditioning and control circuitry selectively enables the second local oscillator when the X, K and K a  radar bands are swept and disables the second local oscillator and bypasses the second mixer when the K u  radar band is swept. 
     The police radar detector may further comprise a first intermediate frequency amplifier passing signals encompassing the second intermediate frequency and amplifying the first intermediate frequency signals. For this embodiment, the first intermediate frequency amplifier couples the first mixer to the second mixer which then mixes amplified first intermediate frequency signals with the second local oscillator signal. When the second intermediate frequency is about 725 megahertz, the first range of frequencies comprises about 14.310 gigahertz to about 15.160 gigahertz. However, the first range of frequencies may comprise about 14.310 gigahertz to about 15.160 gigahertz for the K and K a  radar bands and a subrange of frequencies comprising about 15.090 gigahertz to about 15.160 gigahertz for the X radar band. The second range of frequencies comprises about 14.125 gigahertz to about 14.225 gigahertz. The second mixer may comprise a 90° hybrid circuit and first and second diodes. For this embodiment, one of the first and second diodes is forward biased by the signal conditioning and control circuitry to bypass the second mixer when the K u  radar band is swept. 
     The police radar detector preferably further comprises sideband suppression circuitry for selecting an upper sideband signal or a lower sideband signal from the first intermediate frequency signal when the detector is sweeping the X, K and K a  radar bands. For example, when the second intermediate frequency is about 725 megahertz, the upper sideband signal is around 6.050 gigahertz and the lower sideband signal is around 4.600 gigahertz. The sideband suppression circuitry may comprise a varactor controlled by the signal conditioning and control circuitry to select the upper sideband or the lower sideband. 
     The police radar detector may further comprise a noise suppression circuit which suppresses noise around the second intermediate frequency, for example around 725 megahertz, when the X, K and K a  radar bands are swept. When a noise suppression circuit is provided, it may comprise a diode coupled between the signal conditioning and control circuitry and an input of the second mixer stage, the diode being forward biased when the X, K and K a  radar bands are swept and being reversed biased when the K a  radar band is swept. 
     In accordance with another aspect of the present invention, a police radar detector comprises an antenna for receiving incoming electromagnetic signals. A first local oscillator generates a first local oscillator signal which is swept through a range of frequencies to sweep the X, K and K a  radar bands. A first mixer is coupled to the antenna and the first local oscillator for mixing the incoming electromagnetic signals with the first local oscillator signal to generate first intermediate frequency signals. A second local oscillator generates a second local oscillator signal. A second mixer is coupled to the first mixer for mixing first intermediate frequency signals with the second local oscillator signal to generate second intermediate frequency signals at a second intermediate frequency. Detector circuitry is coupled to the second mixer for detecting received electromagnetic signals within the X, K and K a  bands. Sideband suppression circuitry suppresses an upper sideband signal or a lower sideband signal from the first intermediate frequency signal when the detector is sweeping the X, K and K a  radar bands. Signal conditioning and control circuitry selectively enables the sideband suppression circuitry for selectively enabling suppression of the upper sideband or the lower sideband. 
     The police radar detector may further comprise a first intermediate frequency amplifier passing signals encompassing the second intermediate frequency and amplifying the first intermediate frequency signals, the first intermediate frequency amplifier coupling the first mixer to the second mixer which then mixes amplified first intermediate frequency signals with the second local oscillator signal. When the second intermediate frequency is about 725 megahertz, the upper sideband signal is around 6.050 gigahertz and the lower sideband signal is around 4.600 gigahertz. Also, the range of frequencies comprises about 14.310 gigahertz to about 15.160 gigahertz. However, the range of frequencies may comprise about 14.310 gigahertz to 15.160 gigahertz for the K and K a  radar bands and a subrange of the range of frequencies comprising about 15.090 gigahertz to about 15.160 gigahertz for the X radar band. 
     In the police radar detector, the range of frequencies used to sweep the X, K and K a  bands may comprise a first range of frequencies with the first local oscillator being swept through a second range of frequencies to sweep the K u  radar band, the first intermediate frequency amplifier encompasses the second intermediate frequency and the detector circuitry further detects received electromagnetic signals within the K u  radar band. The signal conditioning and control circuitry further selectively enables the second local oscillator when the X, K and K a  bands are swept and disables the second local oscillator and bypasses the second mixer when the K u  band is swept and wherein the second mixer comprises a 90° hybrid circuit and first and second diodes with one of the first and second diodes being forward biased by the signal conditioning and control circuitry to bypass the second mixer when the K u  band is swept. When the second intermediate frequency is about 725 megahertz, the upper sideband signal is around 6.050 gigahertz and the lower sideband signal is around 4.600 gigahertz. The sideband suppression circuitry may comprise a varactor controlled by the signal conditioning and control circuitry to select the upper sideband or the lower sideband. 
     It is, thus, an object of the present invention to provide a novel frequency scheme for a police radar detector; to provide a novel frequency scheme for a police radar detector which enables the detector to scan the X, K, K u  and K a  radar bands; to provide a novel frequency scheme for a police radar detector which detects the X, K and K a  radar bands using two initial frequency conversions and the K u  radar band using a single initial frequency conversion performed by disabling the second mixer; and, to provide a novel frequency scheme for a police radar detector which detects the X, K and K a  radar bands and selectively suppresses an upper sideband or a lower sideband. 
     Other objects and advantages of the invention will be apparent from the following description, the accompanying drawings and the appended claims. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a schematic block diagram of a police radar detector for practicing the invention of the present application; 
     FIG. 2 is a schematic block diagram illustrating quadrature correlation for the resolution of frequency ambiguities in the police radar detector of FIG. 1; and 
     FIG. 3 is a schematic block diagram of the second mixer stage of the police radar detector of FIG.  1 . 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Reference will now be made to FIG. 1 which illustrates, in schematic block diagram form, a police radar detector  100  utilizing heterodyne receiver principles and being suitable for practicing the invention of the present application. The police radar detector  100  includes two input stages  102 ,  104  which are substantially the same so that only the input stage  102  will be described herein. The input stages  102 ,  104  correspond to a front channel and a rear channel, respectively. Operation of the radar detector  100  using the two input stages  102 ,  104 , or the front and rear channels, is for detecting police radar signals and determining the directional origin of the signals as taught in U.S. Pat. No. 5,083,129 which is assigned to the same assignee as the present application and is incorporated herein by reference. 
     The input stage  102  includes an antenna  106 , such as a microwave hom, for receiving incoming electromagnetic signals. The antenna  106  is coupled to a first mixer  108  and preamplification circuitry comprising an X band and K u  band preamplifier  110  by a diplexer  112 . In the illustrated embodiment of the police radar detector  100 , the diplexer  112  comprises a K band and K a  band rejection filter  113  which rejects frequencies in a band extending from approximately 24 gigahertz (Ghz) to approximately 36 Ghz. The first mixer  108  as illustrated comprises a pair of antiparallel diodes having a near side  108 A coupled to the antenna  106  by the diplexer  112  and a high pass filter  119  and a far side  108 B coupled to a first intermediate frequency (IF) amplifier  114  through a passive multiplexer  116  and a first IF preamplifier  118 . The first IF preamplifier  118  and first IF amplifier  114  pass signals low enough to encompass the second intermediate frequency, for the illustrated embodiment and hereinafter about 725 megahertz (Mhz). It is noted that the first IF amplifier  114  can be omitted from the police radar detector although some sensitivity would be lost. Also, the first IF amplifier  114  can be bypassed or a separate device handling the second intermediate frequency can be provided. These modifications are considered to be within the scope of the present invention. 
     The passive multiplexer  116  comprises: the high pass filter  119  coupled between the antenna  106  and the first mixer  108  for passing signals in the K band, i.e., from approximately 24.05 Ghz to approximately 24.25 Ghz, and the K a  band, i.e., from approximately 33.40 Ghz to approximately 36.00 Ghz, to the near side  108 A of the first mixer  108 ; a first bandpass filter  120  which is designed to pass frequencies in the X band, i.e., from approximately 10.50 Ghz to approximately 10.55 Ghz, and the K u  band, i.e., from approximately 13.40 Ghz to approximately 13.50 Ghz, to the far side  108 B of the mixer  108 ; and, a local oscillator or second bandpass filter  122  which is designed to pass first local oscillator (LO) signals to the far side  108 B of the first mixer  108 . 
     The first LO signals are generated by a first local oscillator (LO)  124  and amplified by an amplifier  126 . In the illustrated embodiment, the first LO  124  comprises a voltage controlled oscillator (VCO) and generates signals which sweep in frequency from approximately 15.160 Ghz to approximately 14.310 Ghz (alternately from 15.160 Ghz to approximately 15.090 Ghz) for sweeping the X band; from approximately 15.160 Ghz to approximately 14.310 Ghz (effectively approximately 30.320 Ghz to 28.620 Ghz) for simultaneously sweeping portions of the K band and the K a  band; and, approximately 14.225 Ghz to approximately 14.125 Ghz for sweeping the K u  band. The high pass filter  119  substantially prevents LO signals from being passed to and broadcast from the antenna  106  by serving as a short to ground for the LO signals so that the LO drive is delivered substantially to the mixer diodes with relatively little LO power being delivered to the antenna  106 . The passive multiplexer  116  also comprises a low pass filter  128  which is designed to pass frequencies from direct current (dc) to approximately 6 Ghz. The band reject filter  113 , which forms the diplexer  112 , can also be considered as forming part of the passive multiplexer  116  since the passive multiplexer  116  must perform the functions of interconnecting the antenna  106 , the first LO  124 , the first mixer  108  and the first IF amplifier  114  with minimal loss from each source of signals to its respective destination. 
     In the input stage topology, illustrated by the input stage  102 , including the connection of the X band and K u  band preamplifier  110  to the far side  108 B of the mixer  108 , the mixer  108  provides additional attenuation of any signals which may feedback toward the input of the X band and K u  band preamplifier  110 . Attenuation of these feedback signals reduces the possibility of oscillation of the X band and K u  band preamplifier  110 . The input stage topology all provides a very direct and low loss path for K band and K a  band signals from the antenna  106  to the near side  108 A of the mixer  108  which is believed to result in a favorable noise figure on the K and K a  band. More conventional input stage connections recombine preamplified X band signals with a passive K band and K a  band path and apply the recombined signals to a single terminal of the mixer. Such input stages or networks are believed to have higher insertion loss on the K and K a  bands which leads to a commensurate increase in noise figure. 
     Signals from the first IF amplifier  114  are passed to a second mixer stage  200  via a conductor  114 A, see FIG.  3 . The second mixer stage  200  comprises a 900° hybrid circuit  202 , a first diode  204  and a second diode  206  which together serve as a second mixer. The circuit  202  receives second LO signals generated by a second LO  208  having a frequency of approximately 5.325 Ghz in the illustrated embodiment when the second LO is active. Signals from the circuit  202  are passed to a second IF amplifier  210  and from there to a bandpass filter  136  via a conductor  136 A. In the illustrated embodiment, both the second IF amplifier  210  and the bandpass filter  136  have a frequency of around 725 Mhz. The bandpass filter  136  is connected to a single sideband down converter  137  comprising a third  138  and a third LO  140  operating at about 725.3 Mhz to down convert signals to approximately 300 kilohertz (Khz). The single sideband down conversion avoids degradation of the system noise figure by approximately 3 dB. The circuitry beyond the second mixer forms detector circuitry for detecting electromagnetic signals in the scanned radar bands. 
     A 300 Khz bandpass filter  142  passes signals from the down converter  137  to an amplitude detector and frequency modulation (FM) detector circuit  144 , sometimes referred to as a discriminator or demodulator circuit, which operates in a manner disclosed in U.S. Pat. No. 5,068,663, which is assigned to the same assignee as the present application and is incorporated herein by reference. The FM output or frequency demodulation detection signal from the circuit  144  is passed to the third LO  140  to lock the radar detector  100  onto incoming electromagnetic signals and also to signal conditioning and control circuitry  146  which includes a processor, preferably any one of a number of appropriate microprocessors. 
     Modulation circuitry  148 , comprising a summer  150  and a 90° phase shifter circuit  152 , is coupled between the signal conditioning and control circuitry  146   6 and the first LO  124  to connect a first modulation signal to the first local oscillator  124 . A second modulation signal, generated by the signal conditioning and control circuitry  146 , is connected directly to the second local oscillator  208  via a conductor  151 . Upon detecting a valid radar signal, i.e., a radar signal in one of the X, K u , K or K a  bands, the signal conditioning and control circuitry  146  activates alarm circuits  154  which can be audible, visual including numeric, directional arrows, or other appropriate for a given alerting arrangement in a known manner. 
     With the understanding of the radar detector  100  gained from the above overview, various aspects of the radar detector  100  will now be described in more detail. The frequencies that are scanned to cover the bands of interest are: the X band —10.50 Ghz to 10.55 Ghz; the K u  band —13.40 Ghz to 13.50 Ghz; the K band—24.05 Ghz to 24.25 Ghz; and, the K a  band—33.40 Ghz to 36.00 Ghz. The frequency scheme of the radar detector  100 , i.e., the frequencies used within the police radar detector including frequencies or swept frequency bands of local oscillators and frequencies of intermediate amplifiers and the tuning methods, provides for sweeping portions of first and second police radar bands, the K band and the K a  band, during a single sweep of the first LO signal generated by the first LO  124  under the control of the signal conditioning and control circuitry  146 . 
     In particular, a 725 MHz second IF amplifier 210 in conjunction with the 5.325 Ghz second LO  208  lead to potential reception of signals at 4.600 Ghz and 6.050 Ghz, i.e., 5.325 Ghz±725 Mhz, in the first IF amplifier  114  and the first IF preamplifier  118 . In the present invention, one of these responses is suppressed so that the radar detectors noise figure is not degraded. Each of these two IF responses, 4.600 Ghz and 6.050 Ghz, combine with the first LO signals from the first LO  124  to yield a total of four receive frequencies, i.e., first LO±4.600 Ghz and first LO±6.050 Ghz. Sideband suppression circuitry comprises a varactor  212  which is tuned by a floating stub  214  and a dc bias or signal suppression signal applied by the signal conditioning and control circuitry  146  via a conductor  218  through a resistor  216  to select the desired sideband and suppress noise in the undesired image sideband. Thus, through sideband suppression, the 4.600 Ghz or 6.050 Ghz response is selected such that two sweeps of the first LO  124  are required to scan the K band and the K a  band. 
     Reference should now be made to Table 1 which details the frequencies adopted in the illustrated embodiment of the radar detector  100 . Each row in Table 1 is labeled with a reference number, i.e., rows 1-10, in the first column. In rows 1-4, the first LO  124  signal (Fvco) is swept from approximately 15.160 Ghz to approximately 14.310 Ghz as shown in the second column of Table 1. But, in the harmonic operating mode of the first mixer  108  of rows 1-4, the local oscillator frequency is inherently doubled to yield injection of first local oscillator signals (Fl) having frequencies which are swept from 30.320 Ghz to 28.620 Ghz as shown in column three. The frequency doubling is due to the antiparallel diodes of the first mixer  108  conducting on alternate half-cycles of the LO signal from the first LO  124  in the harmonic operating mode. In rows 5-8, the first LO  124  signal (Fvco) preferably is swept from approximately 15.160 Ghz to approximately 15.090 Ghz; however, a sweep from approximately 15.160 Ghz to approximately 14.310 Ghz may also be used. 
     The fourth column in each row shows the four different functions or equations which describe frequency conversion from the frequency of received electromagnetic signals (Frf) to the output (F3) of the second IF amplifier  210  at approximately 725 Mhz. The fifth column lists the corresponding frequency ranges of incoming electromagnetic signals Frf that are searched as the first LO  124  is swept across its tuning bandwidth. The sixth or Coverage column of Table 1 shows the particular radar bands or portions of bands which are covered by the receiver responses. The seventh column of Table 1 shows the frequency equations of the third column of Table 1 solved for the second IF amplifier  210  output, F3. These relationships are repeated for convenience in Table 2 which will be referred to later herein. The eighth and ninth columns of Table 1 show the 1st and 2nd intermediate frequencies with the 1st IF frequencies changing between 6.050 Ghz and 4.600 Ghz depending on which sideband is being suppressed. 
     To provide coverage of the X band/K u  band, the operating mode of the first mixer  108  is switched between the K band/K a  band operating mode and the X band/K u  band operating mode by band switching circuitry  157  comprising a resistor  156  and a conductor  158  which extends between the signal conditioning and control circuitry  146  and the X band and K u  band preamplifier  110 . The signal conditioning and control circuitry  146  is thereby able to apply a direct current (dc) bias to the diodes which make up the first mixer  108  through the low pass filter  128 , i.e., to the far side of the first mixer  108 . The dc bias upsets the balance of the diodes which make up the first mixer  108  so that one of the diodes is forward biased and the other diode is reversed biased. 
     This dc bias causes the first mixer  108  to mix more efficiently with the fundamental of the applied first LO signals rather than the second harmonic of the signals generated by the first LO  124 . The dc bias also enables the X band and K u  band preamplifier  110  which is disabled during the K band/K a  band operating mode. 
     The changed operating mode for X band/K u  band reception is indicated in rows 5-10 of Table 1, rows 5-8 for the X band and rows 9-10 for the K u . band. Note that for the X band, four responses again are generated; however, only the response of row 7 provides a signal of interest. Accordingly, the sensitivity of responses in rows 5 and 8 is reduced by suppression of the 6.050 Ghz sideband and, to further reduce noise within he radar detector  100 , the X band and K u  band preamplifier  110  may be designed to uppress the unwanted responses of row 6. 
     For K u  band reception, the mixing operations of the second mixer stage  200  are stopped by disabling the second LO  208  via a conductor  208 A and providing current through a resistor  220  via a conductor  221  to forward bias the diode  206  and thereby bypass the second mixer. The desired signal is then converted directly to 725 Mhz and passed through the forward biased diode  206  to the bandpass filter  136 . The response of row 10 contributes image noise which may be attenuated by selectivity in the X band and K u  band preamplifier  110 . The state of the sideband suppression signal on the conductor  218  is generally irrelevant during the K u  band scanning. It is noted, however, that one or the other of the two states of the sideband suppression circuitry may be preferred dependent on the K u  band tuning characteristics if that state enhances K u  band tuning. 
     During operation of the police radar detector  100  to detect the X, K and K a  bands with the second mixer stage  200  driven by the 5.325 Ghz signal from the second LO  208 , the 725 Mhz response causes a problem. Noise from the front end at 725 Mhz bleeds through the second mixer stage  200  overlaying the desired signals and degrading the noise figure by raising the noise floor. To overcome this problem, a noise suppression circuit comprising a PIN diode  222  is provided. The PIN diode  222  is forward biased through a resistor  224  via a conductor  226  during all modes employing 5.325 Ghz mixing to shunt the725 Mhz noise. During K u  band operation, the diode 222 is reverse biased so that the 725 Mhz signals can be processed. 
     During operation of the police radar detector  100  in the X, K and K a  bands, first IF frequencies are either 4.600 Ghz or 6.050 Ghz. The present invention offers noise figure improvement by alternately suppressing one of these responses while passing the other. For example, if the first IF amplifier  114  delivers output noise power substantially above room temperature (300° K) with equal contributions at 4.600 Ghz and 6.050 Ghz, then a 3 dB improvement accrues by suppressing the undesired sideband. The selected first IF sideband in turn gives rise to two RF responses. Thus, a 6.050 Ghz IF yields two receiver responses about 12 Ghz apart, allowing 24.270 Ghz and 36.370 Ghz to be tuned simultaneously (rows 1 and 4), and a 4.600 Ghz IF yields two receiver responses about 9 Ghz apart, allowing 24.720 Ghz and 34.920 Ghz to be tuned simultaneously (rows 2 and 3). By extension, using two responses, e.g., at 6.050 Ghz and 4.600 Ghz, allows the K a  band to be covered with reduced vco tuning bandwidth. 
     During operation of the police radar detector  100  in the X band, the X band and K u  band preamplifier  110  precedes the first mixer  108  potentially delivering noise at an image frequency that could degrade noise figure. The image occurs at 19.7 Ghz, see row 6 of Table 1, and sideband suppression is again appropriate. The design of the X band and K u  band preamplifier  110  can incorporate selectivity that attenuates the image noise thereby avoiding degraded noise figure. 
     In contrast, there is little benefit to performing single sideband reception forward of the first mixer if the front end is passive, i.e., no preamplifier precedes the first mixer  108 for example, during operation of the police radar detector  100  in the K and K a  bands. There is no significant noise power at the output of the first mixer  108  associated with the undesired sideband, i.e., noise power is near room temperature, such that nothing is gained by suppressing one of the sidebands. Rather, in the police radar detector  100  of the present application, the double sideband front end response is beneficial since it provides coverage of two frequencies of interest simultaneously. System sensitivity improves because, for a given detection bandwidth, the required spectrum can be inspected more frequently offering more opportunity for detection. 
     It is noted that in the process of covering the 2.6 Ghz wide K a  band, the receiver also scans other frequencies where unwanted signals may be generated and these signals must ultimately be ignored. The ability to ignore these unwanted signals is performed by a combination of frequency calibration of the first LO  124 , i.e., the point in the sweep at which a signal is detected is monitored as disclosed in referenced U.S. Pat. No. 5,068,663; and, by being able to identify the active mixer sidebands as will now be described. 
     An important feature of most radar detectors is the ability not only to generate a warning or alert when a radar signal is received, but also to identify the radar band in which the signal originates. When a signal is received in the radar detector  100 , the frequency of the first LO  124  is known because of calibration of the first LO  124 , i.e., the point in the sweep of the first LO  124  at which a signal is detected is noted by the signal conditioning and control circuitry  146  as described above. However, for the X, K and K a  bands, four frequency conversion paths are provided with two being active simultaneously in the police radar detector  100  and the other two being moderately attenuated by sideband suppression (for the K u  band, two frequency conversion paths are active simultaneously) so it is not immediately apparent which receiver response is producing the signal. This ambiguity in the received frequency must be resolved in order to uniquely identify the origin of an incoming signal. 
     The circuit  144  includes an FM detector which generates a frequency demodulation detection signal as its output. If a modulation signal, such as a 500 hertz sine wave, is applied for example to the second LO  210 , the modulation will be superimposed upon the received signal and will appear at the FM detector output  160 . The detected modulation will be either in phase or 180° out of phase with the applied modulation, depending on whether the signal path was the upper sideband or the lower sideband of the second mixer. Analogous results occur if modulation is applied to the first LO  124 . Thus, if a modulation signal, such as a 500 hertz sine wave, is applied to the first LO  124 , the modulation will be superimposed upon the received signal and will appear at the FM detector output  160 . Again, the detected modulation will be either in phase or 180° out of phase with the applied modulation, depending on whether the signal path was the upper sideband or the lower sideband of the first mixer. These operations are illustrated in Table 2 which shows the frequency conversion equations of the fourth column of Table 1 solved for the IF output frequency, F3, i.e., the second IF. 
     Examining the response of row 3, for example, if frequency modulation is applied to the first LO  124 , increasing F 1  produces decreasing F3. The detected frequency modulation would thus be 180° out of phase with the applied modulation. Conversely, modulating the second LO  132  upward in frequency would generate increasing F3. Thus, detected FM would be in phase with modulation applied to F2. Stated differently, the partial derivatives or slopes of F3 with respect to Fl and F2 are −1 and +1, respectively. These slopes for the receiver responses are tabulated in Table 2. Inspection reveals four different combinations for the X, K and K a  bands that in 2-bit binary fashion determine the active signal path. For the K u  band, slope F3/F1 alone determines the active signal path. 
     
       
         
               
             
               
               
               
               
               
               
               
               
               
             
           
               
                 TABLE 1 
               
             
             
               
                   
               
               
                 Receiver Frequency Scheme 
               
             
          
           
               
                   
                   
                   
                 Frf 
                   
                   
                   
                 1st 
                 2nd 
               
               
                 Ref. 
                 Fvco 
                 F1 
                 Function 
                 Frf 
                 Coverage 
                 IF Function 
                 IF 
                 IF 
               
               
                   
               
               
                 1) 
                 15.160 → 14.310 
                 30.320 → 28.620 
                 Frf = 
                 36.370 → 34.670 
                 Top of Ka 
                 F3 = 
                 6.050 
                 0.725 
               
               
                   
                   
                   
                 F1 + F2 + F3 
                   
                   
                 Frf − F1 − F2 
               
               
                 2) 
                 15.160 → 14.310 
                 30.230 → 28.620 
                 Frf = 
                 34.920 → 33.220 
                 Bottom of Ka 
                 F3 = 
                 4.600 
                 0.725 
               
               
                   
                   
                   
                 F1 + F2 − F3 
                   
                   
                 −Frf + F1 + F2 
               
               
                 3) 
                 15.160 → 14.310 
                 30.320 → 28.620 
                 Frf = 
                 25.720 → 24.020 
                 K 
                 F3 = 
                 4.600 
                 0.725 
               
               
                   
                   
                   
                 F1 − F2 + F3 
                   
                   
                 Frf − F1 + F2 
               
               
                 4) 
                 15.160 → 14.310 
                 30.320 → 28.620 
                 Frf = 
                 24.270 → 22.570 
                 Interference 
                 F3 = 
                 6.050 
                 0.725 
               
               
                   
                   
                   
                 F1 − F2 − F3 
                   
                 &amp; K 
                 −Frf + F1 − F2 
               
               
                 5) 
                 15.160 → 15.090 
                 15.160 → 15.090 
                 Frf = 
                 21.210 → 21.140 
                 Of no 
                 F3 = 
                 6.050 
                 0.725 
               
               
                   
                   
                   
                 F1 + F2 + F3 
                   
                 interest 
                 Frf − F1 − F2 
               
               
                 6) 
                 15.160 → 15.090 
                 15.160 → 15.090 
                 Frf = 
                 19.760 → 19.690 
                 Of no 
                 F3 = 
                 4.600 
                 0.725 
               
               
                   
                   
                   
                 F1 + F2 − F3 
                   
                 interest 
                 −Frf + F1 + F2 
               
               
                 7) 
                 15.160 → 15.090 
                 15.160 → 15.090 
                 Frf = 
                 10.560 → 10.490 
                 X 
                 F3 = 
                 4.600 
                 0.725 
               
               
                   
                   
                   
                 F1 − F2 + F3 
                   
                   
                 Frf − F1 + F2 
               
               
                 8) 
                 15.160 → 15.090 
                 15.160 → 15.090 
                 Frf = 
                 9.110 → 9.040 
                 Of no 
                 F3 = 
                 6.050 
                 0.725 
               
               
                   
                   
                   
                 F1 − F2 − F3 
                   
                 interest 
                 −Frf + F1 − F2 
               
               
                 9) 
                 14.225 → 14.125 
                 14.225 → 14.125 
                 Frf = 
                 13.500 → 13.400 
                 Ku 
                 F3 = 
                 0.725 
               
               
                   
                   
                   
                 F1 − F3 
                   
                   
                 −Frf + F1 
               
               
                 10) 
                 14.225 → 14.125 
                 14.225 → 14.125 
                 Frf = 
                 14.950 → 14.850 
                 Of no 
                 F3 = 
                 0.725 
               
               
                   
                   
                   
                 F1 + F3 
                   
                 interest 
                 Frf − F1 
               
               
                   
               
               
                 F1 is the first mixer local oscillator injection and is 2*Fvco in Responses 1-4; F1 = Fvco in responses 5-10.  
               
               
                 F2 is the second mixer local oscillator frequency (5.325 Ghz).  
               
               
                 F3 is the second intermediate amplifier frequency (725 Mhz).  
               
             
          
         
       
     
     
       
         
               
             
               
               
               
               
             
           
               
                 TABLE 2 
               
             
             
               
                   
               
               
                 FM Detection Phase 
               
             
          
           
               
                 Ref. 
                 IF Function 
                 Slope F3/F1 
                 F3/F2 
               
               
                   
               
               
                 1) 
                 F3 = Frf − F1 − F2 
                 −1 
                 −1 
               
               
                 2) 
                 F3 = −Frf + F1 + F2 
                 +1 
                 +1 
               
               
                 3) 
                 F3 = Frf − F1 + F2 
                 −1 
                 +1 
               
               
                 4) 
                 F3 = −Frf + F1 − F2 
                 +1 
                 −1 
               
               
                 5) 
                 F3 = Frf − F1 − F2 
                 −1 
                 −1 
               
               
                 6) 
                 F3 = −Frf + F1 + F2 
                 +1 
                 +1 
               
               
                 7) 
                 F3 = Frf − F1 + F2 
                 −1 
                 +1 
               
               
                 8) 
                 F3 = −Frf + F1 − F2 
                 +1 
                 −1 
               
               
                 9) 
                 F3 = −Frf + F1 
                 +1 
                 0 
               
               
                 10) 
                 F3 = Ffr − F1 
                 −1 
                 0 
               
               
                   
               
             
          
         
       
     
     When a signal is acquired, the signal conditioning and control circuitry  146  of the radar detector  100  performs these operations to identify the active frequency conversion function. Measurement is accomplished in an economical manner by employing orthogonal modulation at the two local oscillators that is analogous to quadrature phase shift keying (QPSK ) used in data communication. Synchronous quadrature correlation of the detected frequency modulation yields either positive or negative correlation with each local oscillator. The correlations correspond directly to the aforementioned slopes, thus resolving the frequency ambiguity. It is therefor apparent that −1, −1 and +1, +1 identify a detected incoming signal as being in the K a  band while −1, +1 and +1, −1 identify a detected incoming signal as being in the K band. When detecting X band, −1, +1 identify a detected incoming signal as being in the X band. When detecting the K u  band, +1 (slope F3/F1) identifies a detected incoming signal as being in the K u  band. 
     A hardware implementation of quadrature correlation is conceptually illustrated in the schematic block diagram of FIG.  2 . The signal conditioning and control circuitry  146  of the police radar detector  100  generates a 500 Hz sine wave which is passed through the modulation circuitry  148  to result in a first modulation signal being applied to the first LO  124  with the 500 Hz signal serving as a second modulation signal which is applied to the second LO  210 . Thus, the first and second modulation signals are in quadrature to one another. 
     Correlated detection is performed by multiplying or mixing the detected FM signal on a conductor  162  with the 500 Hz signal generated by the signal conditioning and control circuitry  146  on a conductor  164  in a mixer circuit  166  and integrating the result over one or more integer periods of the modulation in a first integrator circuit  168 ; and, mixing the detected FM signal on the conductor  162  in a mixer  167  with the 500 Hz signal which has been shifted by −90° on the conductor  169  and integrating the result over one or more integer periods of modulation in a second integrator circuit  170 . The output signals from the integrator circuits  168  and  170  are passed through threshold circuits  172 ,  174 , respectively, to generate the +1 or −1 slopes which are sent to the signal conditioning and control circuitry  146  on conductors  176 ,  178  where the operations described above are performed to determine the radar bands of received electromagnetic signals. 
     It is well known from modem theory that the quadrature modulation arrangement allows two independent data bits to be transmitted simultaneously. In analogous fashion, the quadrature modulation arrangement permits characterization of both frequency conversions simultaneously and operates with an economy of hardware and signal analysis time. While a hardware implementation is shown in FIG. 2, the synchronous quadrature correlation for the radar detector  100  is preferably implemented by digitizing the detected FM signal and performing the multiplications and integrations numerically in software within a microprocessor of the signal conditioning and control circuitry  146 . 
     The quadrature correlation technique is very economical since the 90° phase shifter circuit  152  can be constructed in a conventional manner using only one capacitor and a few resistors and the quadrature detection requires only additional software for performing operations corresponding to the operations performed by the hardware of FIG.  2 . While the above described modulation/quadrature correlation techniques are preferred for the radar detector  100 , it is apparent that other modulation techniques can also be utilized. For example, modulation signals having different frequencies can be applied to one or both mixers. 
     By using the above described techniques, the frequency of any signal detected by the radar detector  100  can be uniquely identified. A review of rows 1-4 of Table 1 shows that some portions of the K band, from 24.020 Ghz to 24.270 Ghz, and some portions of the K a  band, from 34.670 Ghz to 34.920 Ghz are scanned twice because of overlapping frequency coverage. Accordingly, indiscriminate counting of valid K and K a  band responses would lead to some double counting of signals lying within the overlapping regions. To overcome these counting problems, the radar detector  100  is characterized during manufacturing to determine the A/D codes of the sweep signals corresponding to 34.670 Ghz in row 2 and 35.920 Ghz in row 1. The measured codes are stored so that they can be used in connection with band identification to avoid double counting of signals within the overlapping regions. Double counting of K band signals is precluded by simply ignoring K band signals received in the responses of row 4 of Table 1. This signal identification permits determination of the accurate number of radar sources incident on the radar detector  100  so that the user of the radar detector  100  can be notified. Detection and alerting of multiple radar sources is described in U.S. Pat. No. 5,146,226 which is assigned to the same assignee as the present application and is incorporated herein by reference. 
     Additional details regarding operation of the preferred embodiment of the radar detector  100  can be determined by review of U.S. Pat. Nos. 5,852,417; 5,856,80; 5,900,832; and, 5,917,441 which are assigned to the same assignee as the present application and are incorporated herein by reference. 
     Having thus described the invention of the present application in detail and by reference to preferred embodiments thereof, it will be apparent that modifications and variations are possible without departing from the scope of the invention defined in the appended claims. For example, while the second IF is preferably selected as illustrated at 725 Mhz to help minimize the tuning bandwidth required for the first LO, other second IF frequencies can be employed in the present invention.