Abstract:
Embodiments include a method to suppress side lobes of a main antenna by creating cancellation directions using a Side Lobe Canceller (SLC). Various embodiments also provide the Side Lobe Canceller.

Description:
RELATED 
       [0001]    This application claims priority under 35 U.S.C. 119 to European Patent Application No. EPO 08446503.8, filed 7 Feb. 2008, which application is incorporated herein by reference and made a part hereof. 
       TECHNICAL FIELD 
       [0002]    The invention relates to the field of suppression of side lobes of antennas. 
       BACKGROUND ART 
       [0003]    Cancellation of side lobes are often used in radar systems in order to cancel jammer signals picked up in side lobes of a radar antenna. The existing solutions allow jammer signals from several jammer sources to be cancelled. One solution is described in U.S. Pat. No. 4,044,359 “Multiple Intermediate frequency side-lobe canceller”. Side lobes are difficult to avoid when designing antennas and jammer signals can be picked up by these side lobes. As the jammer signals are often powerful they will cause interference with the signal from a target picked up in the main lobe even if the reception sensitivity of the side lobe is much below the reception sensitivity of the main lobe. 
         [0004]    The principle of operation of prior art solutions is shown in  FIG. 1  with a main antenna  101 , an auxiliary array antenna  102  having N antenna elements A 1  to A N , a control unit  103  and a cancelling circuit  104  with an output terminal  111 . The main antenna has a main lobe  104  and side lobes  105 - 108 . Jammer signals are emitted from jammer sources J 1 -J 3 . As can be seen in  FIG. 1  jammer signals from J 1  are picked up by the side lobe  108  in the main antenna and by the antenna elements in the auxiliary antenna  102 . Signals s 1 -s N  from the antenna elements A 1  to A N  are fed to the control unit  103 . The control unit produces error signals e 1 -e N  which are fed to the cancelling circuit  104 . The output from the main antenna  101  is fed to the cancelling circuit  104 . There is also a feedback signal  112  from the output terminal  111  of the cancelling circuit to the control unit  103 . In the control unit the feed back signal will be used to affect amplitude and phase of the input signals s 1 -s N  to produce the error signals e 1 -e N . When the error signals are subtracted in the cancelling circuit with the undesired signal from side lobes of the main antenna they will cancel the jammer signal so that only the signal representing the main lobe and non affected side lobes will be available at the output terminal  111 . Thus the undesired signal/s picked up by the undesired side lobe/s is/are cancelled. 
         [0005]    The drawback with the existing solutions today is that only narrow band cancellation of the side lobe is possible. This is a serious problem as radar antennas are often operating over a very wide bandwidth and the control unit is only capable of producing the error signals over a narrow bandwidth resulting in that the side lobe cancellation is only effective in a part of the operating bandwidth of the radar antenna. 
         [0006]    Thus there is a need for an improved solution for providing a side lobe cancellation over a wide bandwidth in order to cancel jammer interference over a wide bandwidth. 
       SUMMARY OF THE INVENTION 
       [0007]    The object of the invention is to remove the above mentioned deficiencies with prior art solutions and provide:
       a method to suppress side lobes of a main antenna   a Side Lobe Canceller (SLC)
 
to solve the problem to achieve a side lobe cancellation over a wide bandwidth in order to cancel jammer interference over a wide bandwidth.
       
 
         [0010]    This object is achieved by providing a method to suppress side lobes of a main antenna by creating cancellation directions using a Side Lobe Canceller (SLC) comprising an auxiliary antenna and an Adaptive Side lobe Canceller control unit (ASC), the auxiliary antenna having N antenna elements/sub arrays 1 to N, N being ≧1, feeding their output waveform/s s 1  to s N  to the ASC. The SLC further comprises a feedback loop from an output terminal of the SLC to the ASC, wherein:
       the Side Lobe Canceller (SLC) further comprising N transforming means Tr 1  to Tr N      the ASC receives an output waveform from the main antenna   each output waveform s 1  to S N  from the antenna elements of the auxiliary antenna is fed also to the transforming means Tr 1  to Tr N      a delay is introduced in the output waveform through a delay line   a weighting function W(ω) being calculated for Q spectral components q, resulting from dividing the bandwidth B in q components, q being an integer index ranging from 0 to Q−1, for each antenna element or sub array (A 1 -A N ) by minimizing the jammer influence on the output waveform z   the transforming means affecting the waveforms from each antenna element/sub array by use of one or several control parameters calculated from the weighting function W(ω) at discrete angular frequencies ω q      the control parameters being continuously calculated in the ASC and fed to the transforming means through control signals c 1 , to c N  
 
thus achieving cancellation of jammer interference over the entire bandwidth B when the output waveforms from the transforming means are subtracted in the time or frequency domain from the delayed output waveform of the main antenna.
       
 
         [0018]    The object is further achieved by providing a Side Lobe Canceller (SLC) used to suppress side lobes of a main antenna by creating cancellation directions. The SLC comprises an auxiliary antenna and an Adaptive Side lobe Canceller control unit (ASC), the auxiliary antenna having N antenna elements/sub arrays 1 to N, N being ≧1, arranged to feed their output waveform/s s 1  to s N  to the ASC. The SLC further comprises a feedback loop from an output terminal of the SLC to the ASC, wherein:
       the Side Lobe Canceller (SLC) further comprising N transforming means Tr 1  to Tr N      the ASC is arranged to receive an output waveform from the main antenna   each output waveform s 1  to s N  from the antenna elements of the auxiliary antenna is connected also to the transforming means Tr 1  to Tr N      means for a delay in the output waveform is arranged with a delay line   a weighting function W(ω) is arranged to be calculated for Q spectral components q, resulting from dividing the bandwidth B in q components, q being an integer index ranging from 0 to Q−1, for each antenna element or sub array (A 1 -A N ) by minimizing the jammer influence on the output waveform z at a centre frequency f q  of each spectral component and   the transforming means are arranged to affect the waveforms from each antenna element/sub array by use of one or several control parameters calculated from the weighting function W(ω) at discrete angular frequencies ω q      the control parameters are arranged to be continuously calculated in the ASC and to be fed to the transforming means through control signals c 1  to C N  
 
thus achieving cancellation of jammer interference over the entire bandwidth B when the output waveforms from the transforming means are arranged to be subtracted in the time or frequency domain from the delayed output waveform of the main antenna.
       
 
         [0026]    Further advantages are achieved by implementing one or several of the features of the dependent claims which will be explained in the detailed description. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0027]      FIG. 1  schematically shows a prior art solution for side lobe cancellation. 
           [0028]      FIG. 2  shows a block diagram of an embodiment of the invention. 
           [0029]      FIG. 3   a  schematically shows a digital solution of a realization in the frequency domain of a transforming means. 
           [0030]      FIG. 3   b  schematically shows an analogue solution of a realization in the frequency domain of a transforming means. 
           [0031]      FIG. 4   a  schematically shows a realization of a transforming means in the time domain. 
           [0032]      FIG. 4   b  schematically shows a realization in the time domain for an embodiment of the transforming means including also a dominating non frequency dependent “true time delay”. 
           [0033]      FIG. 4   c  shows a diagram of attenuation/amplification and time delays as a function of angular frequency ω(2·π·f). 
           [0034]      FIG. 5  shows the definition of angles φ and θ used in the definition of the antenna pattern. 
           [0035]      FIG. 6  schematically shows radiation patterns of a main antenna and an auxiliary antenna. 
           [0036]      FIG. 7  schematically shows a flow chart of the calculations to obtain the control signals. 
       
    
    
     DETAILED DESCRIPTION 
       [0037]    Embodiments of the invention will now be described in detail with reference to the enclosed drawings. Embodiments of the invention are applicable for all types of receiving antennas. Henceforth in the description the invention will be exemplified with a radar antenna application. 
         [0038]    A block diagram of an embodiment of the invention implemented on a radar system is shown in  FIG. 2 . The main parts of a Side Lobe Canceller (SLC),  200 , shown in  FIG. 2  are a main antenna  201 , an auxiliary antenna  202  consisting of antenna elements or a combination of antenna elements and sub arrays A 1 -A N , an Adaptive Side Lobe Canceller control unit (ASC)  203 , transforming means Tr 1  to Tr N , a Cancellation Network (CN)  204  and a delay line  212 . The number of antenna elements/sub arrays is at least one, i.e. N≧1. A sub array consists of at least two antenna elements. The delay line is inserted between the main antenna and the CN, producing a delayed output waveform  211  of the main antenna. The delay can be adjusted from zero delay and upwards depending on distance between the auxiliary antenna and the main antenna. The delay is accomplished by conventional means well known by the skilled person. The transforming means are affecting waveforms from the auxiliary antenna. Each antenna element in the auxiliary antenna is connected to a transforming means in the embodiment of  FIG. 2 . Before  FIG. 2  is explained in detail the transforming means will be further described. Henceforth in the description the term waveform is used to designate a signal in any form, continuous or pulsed. 
         [0039]    The transforming means in the embodiment of  FIG. 2  will affect input waveforms s 1 -s N  entering the transforming means such that output waveforms s 1 ′ to s N ′ from the transforming means, when each of them is subtracted from the delayed output waveform  211  of the main antenna, can at most produce one cancellation direction for each waveform s 1 ′ to s N ′. Alternatively the waveforms s 1 ′ to s N ′ are first summarized in a first summation network  206  and the summarized signal is subtracted from the delayed output waveform  211  of the main antenna. Within the transforming means the waveforms will be affected by certain control parameters, continuously updated by the ASC  203  and explained further below.  FIG. 3   a  schematically shows an example of a practical realization of a transforming means called a first control element. In this example a time delay being frequency dependent is used as a control parameter. The realization also includes another control parameter being a frequency dependent attenuation/amplification, i.e. the amplitudes of the waveforms are attenuated or amplified. In this embodiment two frequency dependent control parameters thus are used; time delay and attenuation/amplification. An input waveform s in (t),  301 , representing one of the input waveforms s 1  to s N , from an antenna element n in the auxiliary antenna, is fed to a Fourier Transformation (FT) unit  302  using for example a Fast Fourier Transformation (FFT), but other methods for calculation of the spectrum can be used. The FT unit transforms the input waveform s in (t),  301 , into Q spectral components 0 to Q−1, in this example into 8 spectral components  310 - 317 , each spectral component having a centre frequency f q . However the transformation can be made into more or less spectral components. This means that a total bandwidth B covered by the radar system is divided in Q spectral components. The time delay τ n,q , ( 320 - 327 ) and the optional frequency dependent attenuation/amplification a n,q  ( 330 - 337 ) are affecting each spectral component q from auxiliary antenna element n through any suitable time delay and/or attenuation/amplification means well known to the skilled person. The spectral component  310  thus has a time delay τ n,0 ,  320 , and an attenuation/amplification a n,0 ,  330 , the spectral component  311  a time delay τ n,1 ,  321 , and an attenuation/amplification a n,1 ,  331 , and so on until the spectral component  317  having a time delay τ n,7 ,  327 , and an attenuation/amplification a n,7 ,  337  in auxiliary antenna element n. All spectral components are fed to an Inverse Fourier Transformation (IFT) unit,  303 , using Inverse Fast Fourier Transformation (IFFT) or any other suitable transformation method, as for example IDFT, transforming from the frequency domain to the time domain thus transforming all the spectral components back into the time domain and producing an output waveform s out (t),  304 , representing one of the output waveforms s 1 ′ to S N ′. 
         [0040]    The time delay τ n,q  and the attenuation/amplification a n,q  are examples of control parameters for auxiliary antenna element n affecting each spectral component q where n ranges from 1 to N and q from 0 to Q−1. 
         [0041]    The FT unit, the time delay and attenuation/amplification means and the IFT unit are parts of the first control element  300 . 
         [0042]    The function of the implementation with both the frequency dependent time delay and the attenuation/amplification according to  FIG. 3   a  will now be described. 
         [0043]    The control parameters mentioned above are calculated from a frequency dependent weighting function W(ω)=A(ω)·e −j·ω·τ(ω)  and are affecting the waveforms s 1  to s N  where A(ω), accounts for the frequency dependency of the attenuation/amplification and τ(ω) account for the frequency dependency of the time delay. As an alternative the weighting function could be defined as W(ω)=A(ω)·e −j·φ(ω)  where A(ω), still accounts for the frequency dependency of the attenuation/amplification and φ(ω) accounts for the frequency dependency of the phase shift. Each auxiliary antenna element is connected to one first control element  300 . The output waveform s out (t)  304  emitted from each first control element  300  as a function of the input waveform s in (t)  301  entering the first control element can be calculated with the aid of equation (1). The waveform s in (t) is on video-, intermediate frequency- (IF) or radio frequency (RF)-level. 
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         [0044]    In equation (1) the symbol           symbolize convolution. The principle of convolution is well known to the skilled person and can be further studied e.g. in “The Fourier Transform and its Applications”, McGraw-Hill Higher Education, 1965 written by Ronald N. Bracewell. 
         [0045]    The symbols used above and henceforth in the description have the following meaning:
   ω=angular frequency (2·π·f)   ω q =discrete angular frequency at frequency f q      w(t)=time domain weighting function   W n,q =time domain weighting coefficient equal to component q of the IFT of row n in W   w=Matrix consisting of the weighting coefficients W n,q  (nε[1 . . . N], qε[0 . . . Q−1]) with a row for each antenna element and a column for each spectral component   w(t−τ)=time delayed time domain weighting function   W(ω)=frequency domain weighting function being the Fourier Transform of w(t)   W n,q =W(ω) at ω=ω q  corresponding to spectral component q in auxiliary antenna element n   W=Matrix consisting of the elements W n,q  (nε[1 . . . N], qε[0 . . . Q−1])   A(ω)=absolute value of W(ω)   a n,q =A(ω) at ω=ω q  corresponding to the absolute value of spectral component q in auxiliary antenna element n equal to the absolute value of W n,q      τ=time delay and integration variable   τ(ω)=time delay as a function of ω   τ n,q =time delay of τ(ω) at ω=ω q  corresponding to spectral component q in auxiliary antenna element n   φ(ω)=phase shift as a function of ω   φ n,q =phase shift of φ(ω) at ω=ω q  corresponding to spectral component q in auxiliary antenna element n   S Mq =Spectral component q from the main antenna   S M =Row vector consisting of the elements S Mq (qε[ 0  . . . Q−1]) being the FT of the output waveform of the main antenna   S n,q =Spectral component q from the auxiliary antenna n   S=Matrix consisting of the elements S n,q (nε[1 . . . N], qε[0 . . . Q−1])   P q =Residual power in spectral component q of the output waveform z   P=Row vector consisting of the elements P q (qε[ 0  . . . Q−1])   Z q =Spectral component q of the output waveform z   Z=Vector consisting of the elements Z q  (qε[ 0  . . . Q−1])   z=Output waveform in the time domain   
 
         [0071]    The weighting elements and weighting coefficients mentioned above are examples of control parameters. 
         [0072]    Vectors and matrixes are designated with bold characters. 
         [0073]    As mentioned above τ n,q  and a n,q  are examples of frequency dependent control parameters for auxiliary antenna element n affecting each spectral component q. The phase shift φ n,q  is another example of a frequency dependent parameter for auxiliary antenna element n affecting each spectral component q. 
         [0074]    This embodiment clarifies an important difference between the inventive solution and prior art. In prior art solutions the total output waveform from each antenna element in the auxiliary antenna is treated by affecting phase and amplitude. This treatment is the same over the entire frequency range, i.e. it is frequency independent. In this embodiment of the inventive solution each waveform from the auxiliary antenna elements is divided into a number of spectral components and each spectral component is affected by one or several control parameters depending on frequency. This makes it possible to accurately control gain and delay or phase of each spectral component and thus also to accurately control gain and delay or phase over the total bandwidth B. 
         [0075]      FIG. 3   a  describes a digital realization of the transforming means.  FIG. 3   b  shows a corresponding analogue realization with the input waveform s in (t)  301  entering a third control element  350 . The input waveform  301  coming from auxiliary antenna element n is fed to Q band pass filters F q  having a centre frequency f q  where q assumes integer values from 0 to Q−1. The input waveform  301  is thus split in Q spectral components and a time delay τ n,q  or alternatively a phase shift φ n,q  and a frequency dependent attenuation/amplification a n,q  are affecting each spectral component through any suitable time delay or phase shift and attenuation/amplification means well known to the skilled person. All spectral components are connected to a second summation network  351  producing the output waveform s out (t),  304 . The centre frequency f q  of each spectral component can be calculated according to: 
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         [0000]    for a case with equividistant spectral component division, where f c  is the centre frequency in the total frequency band and B is the total bandwidth. 
         [0076]    The third control element  350  comprises Q band pass filters F q , means for time delay and amplification/attenuation as well as the second summation network  351 . 
         [0077]    A further digital realization of the transforming means will now be described with reference to  FIGS. 4   a  and  4   b . In many situations a time discrete realization, with discrete steps T in time, might be preferable. An output waveform s out (m·T), representing one of the output waveforms s 1 ′ to s N ′, emitted from a second control element ( 400 ) can be calculated with the aid of equation (2) as a function of an input waveform s in (m·T), representing one of the input waveforms s 1  to s N , entering the second control element. The index m is an integer value increasing linearly as a function of time. W(ω q ) represents the time delay and attenuation/amplification at the centre frequency of spectral component q, see  FIG. 3 . The FFT and the IFFT described in association with  FIG. 3   a , both requiring Q·log 2 (Q) operations, are computational efficient methods for calculation of DFT (Discrete Fourier Transform) and IDFT (Inverse Discrete Fourier Transform), both requiring Q 2  operations. Q is as mentioned above the total number of spectral components. The output waveform is calculated as: 
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         mod [x,y]=remainder after division of x by y 
         ω q =2·90 ·f q =discrete angular frequency 
         Q=Number of spectral components 
         k=integer raising variable used in the DFT and the IDFT 
         m=integer raising variable for discrete time steps 
         q=integer raising variable for spectral components and integer raising variable used in the DFT. 
         W(ω q )=frequency domain weighting function for frequency f q  being the Fourier Transform of W n,q  for auxiliary antenna element n. 
       
     
         [0085]    As can be seen in equation (2) the desired functionality in a time discrete realization can be achieved with Q operations. 
         [0086]    FFT and DFT are different methods for Fourier Transformation (FT). IFFT and IDFT are corresponding methods for Inverse Fourier Transformation (IFT). As described above these methods have different advantages and the method most suitable for the application is selected. However any of the methods can be used when FT and/or IFT are/is required in the different embodiments of the invention. 
         [0087]      FIG. 4   a  shows the input waveform s in (m·T)  401 , coming from an antenna element in the auxiliary antenna. The input waveform  401  is successively time delayed in Q−1 time steps T,  403 , numbered from 1 to Q−1 and being time delayed copies of the input waveform s in (m·T). The input waveform is thus successively time delayed with time steps T as illustrated in the upper part,  404 , of  FIG. 4   a . Q control parameters comprising weighting coefficients w n,0  to w n,Q-1 , for auxiliary antenna element n is identified with two indexes, the first representing auxiliary antenna element number and the second a consecutive number q representing a spectral component and ranging from 0 to Q−1. The weighting coefficients are calculated for each antenna element n as the IDFT of W(ω q ) or alternatively as the IFFT of W(ω q ) for the Q spectral components q, resulting from dividing the bandwidth B in q components. The weighting coefficients in the time domain w n,0  to w n,Q-1  thus is the weighting coefficient for antenna element n in the auxiliary antenna. The corresponding coefficients in the frequency domain are designated W n,q . The arrows  411  illustrate that the input waveform s in (m·T) is multiplied with the first weighting coefficient w n,0  and each time delayed copy of the input waveform is successively multiplied with the weighting coefficient having the same second index as the number of time step delays T included in the in the time delayed copy of the input waveform as illustrated in the middle part,  405 , of  FIG. 4   a . The result of each multiplication is schematically illustrated to be moved, indicated with arrows  412 , to the bottom part,  406 , of  FIG. 4   a , where each multiplication result is summarized to the output waveform  407 , s out (m·T). 
         [0088]    If the dominating part of the time delay is not frequency dependent, which is often the case when there is a physical distance between the main antenna and the auxiliary antenna, it will result in many very small consecutive weighting coefficients, approximately equal to zero, at the beginning and end of the series of weighting coefficient w n,0  to w n,Q-1  for each antenna element in the auxiliary antenna. Assume that the first x weighting coefficients and the last y weighting coefficients in the series of weighting coefficients w n,0  to w n,Q-1  are approximately equal to zero. It could then be suitable in a hardware realization, to set the first x weighting coefficients and the last y weighting coefficients to zero and to integrate the first x time delays T into a time delay D,  402 , equal to x·T as illustrated in  FIG. 4   b , and to exclude the last y multiplications to reduce the number of required operations to less than Q operations.  FIG. 4   b  otherwise corresponds to  FIG. 4   a . The time delay D,  402 , corresponds to the non frequency dependent time delay, for each antenna element in the auxiliary antenna. The remaining frequency dependent time delay will onwards be called “delta time delay”.  FIG. 4   b  is an example of a computational efficient convolution, for calculation of the “delta time delay”, preceded of the frequency independent time delay D,  402 , used mainly to compensate for the physical distance between the main antenna and the auxiliary antenna. Compensation for this physical distance is made relative to the non frequency dependent delay line,  212 . The delay time of the non frequency dependent delay line,  212  must correspond to the maximum time difference or delay of the jamming signals in the main antenna and the auxiliary antenna caused by the physical distance between the main antenna and the auxiliary antenna. 
         [0089]    The means for realizing the frequency independent time delay D and the means for frequency dependent time delays and attenuations/amplifications are parts of the second control element  400 . 
         [0090]    The first  300 , second  400  and third  350  control elements are examples of transforming means. In an alternative embodiment with a common IFT, which will be explained below, the IFT part does not have to be included in the digital transforming means and the second summation network  351  does not have to be included in the third control element. 
         [0091]      FIG. 4   c  shows the angular frequency dependency of the time delay τ(ω) and attenuation A(ω) on the vertical axis  415  as a function of the angular frequency ω (i.e. 2·π·f) on the horizontal axis  416 . The optimum weighting function is estimated for each auxiliary antenna element n and for a number of ω-values, ω 0 , ω 1 , ω 2  . . . ω Q-1  for example by minimizing the output residual power from the Cancellation Network (CN) for each spectral component q. The output waveform z ( 207 ) is the difference between the delayed output waveform ( 211 ) from the main antenna and the sum of the waveforms s 1 ′ to s N ′ ( 209 ) when the subtraction is made in the time domain. All jammer signals as well as the useful signal will be present in both the waveform from the main antenna and the waveforms s 1  to s N  from the elements in the auxiliary antenna with varying time delays and strengths. By minimizing the output residual power from the Cancellation Network (CN) for each spectral component q the jammer influence on the output waveform z ( 207 ) will be minimized. To simplify the measurement of the residual power for each spectral component the first summation network and the Cancelling Network could be duplicated for each spectral component thus using Q single spectral first summation networks and Q single spectral Cancellation Networks. This results in one feedback waveform for each spectral component fed to the ASC. The output waveform z,  207 , is then formed as the IFT of all spectral components after the single spectral Cancellation Networks. 
         [0092]    The described calculation is performed for each spectral component q. This results in a number of values W n,0 , W n,1 , W n,2  . . . W n,Q-1  for each auxiliary antenna element n. The time delay as a function of a) then forms a curve  417  and the attenuation/amplification a curve  418 . The weighting coefficients W n,0 , W n,1 , W n,2  . . . W n,Q-1  are calculated as the IDFT or IFFT of W n,0 , W n,1 , W n,2  . . . W n,Q-1  for each auxiliary antenna element n. 
         [0093]      FIGS. 4   a  and  4   b  thus shows a realization of a frequency dependent time delay and attenuation/amplification in the time domain and  FIGS. 3   a  and  3   b  shows a corresponding realization in the frequency domain. An advantage with the realization in the time domain is that only Q operations are required while the realization in the frequency domain requires Q·log 2 (Q) operations as described above. 
         [0094]    All three control elements could as mentioned earlier be inserted either at video, intermediate frequency (IF) or directly on radio frequency (RF) level. It is easier to realize the control element at lower frequency but all hardware needed between the control element and the antenna element/sub array in the auxiliary antenna need to be multiplied with the number of antenna elements/sub arrays in the auxiliary antenna. It should also be noted that the Cancellation Network (CN) and all hardware in the SLC must have a dynamic range wide enough to handle the jammer influence of the received waveform. In the description the invention is henceforth described as being realized at the RF level. 
         [0095]    The three control elements are examples of transforming means, transforming an input waveform to an output waveform. The transforming means all have two ends, an input end receiving the input waveform and an output end producing the output waveform. 
         [0096]    Returning to  FIG. 2 , the Side Lobe Canceller (SLC)  200  will now be described in more detail. Each antenna element A 1 -A N  in the auxiliary antenna is in principal used to cancel the side lobe interference in a certain direction as will be explained below. The auxiliary antenna can be part of the main antenna or separated from the main antenna. The main antenna can be an array antenna with a number of antenna elements or any other type of antenna. Jammer signals are emitted from jammer sources J 1  and J 2  and picked up by the main antenna and the auxiliary antenna. The waveform  205  from the main antenna is fed both to the Adaptive Side lobe Canceller (ASC) control unit  203  and to a Cancellation Network  204  via the delay line  212 . The waveforms s 1  to s N  from each antenna elements 1 to N in the auxiliary antenna are fed to the Adaptive Side lobe Canceller (ASC) control unit  203 . The waveform s 1  is also fed to transforming means unit  1 , Tr 1 , the waveform s 2  to transforming means unit  2 , Tr 2 , and the waveform s n  to transforming means unit n, Tr n , i.e. each waveform s 1  to s N  is fed to corresponding transforming means unit Tr 1  to Tr N . The optimum control parameters W could be estimated by minimizing the residual power in each spectral component q of the difference between each spectral component of the delayed waveform from the main antenna S Mq  and the sum of spectral component q from each output waveform s 1 ′ to s N ′. The output residual power P can be calculated as: 
         [0000]        P=E[|Z|   2   ]=E[|S   M   T   −W   T   −S|   2 ] 
       Where: 
       [0097]    E[x]=Vector containing the expected value of each element in the vector x
 
|x|=Vector containing the absolute value of each element in the vector x
 
X T =Transponat of the matrix X
 
In the second equality:
 
         [0000]        E[|Z|   2   ]=E[|S   M   T   −W   T   ·S|   2 ] 
         [0000]    only the weighting elements W are unknown. This relation together with minimization of E[|Z| 2 ] for all spectral components q can thus be used for calculation of W. When the total residual power P for all spectral components of the output waveform z is minimized the influence of the jammer signal/signals is also minimized, in all practical situations when cancellation is required. The signal to noise ratio of the output waveform z is, during these circumstances, maximized by the operations described above. 
         [0098]    Control parameters equal to row  1  to row N of W or row  1  to row N of w depending on the chosen realization of the transforming means as explained above, is sent to corresponding transforming means Tr 1  to Tr N  via control signals c 1  to c N . In the transforming means the control parameters are received and implemented in means for achieving the frequency dependent time delay or frequency dependent phase shift and the frequency dependent amplification/attenuation when the invention is implemented in the frequency domain or in the means for achieving the multiplication with the weighting coefficients w n,0  to w n,Q-1  when the invention is implemented in the time domain. w n,q  is the weighting function for antenna element n for delay q used when the invention is implemented in the time domain and W n,q  is equal to component q of the FT of row n in w used when the invention is implemented in the frequency domain as explained above. The outputs s 1 ′ to s N ′ from all transforming means, one or several, are added together in the first summation network  206  creating the summarized output waveform  209 . The output waveform  209  from the first summation network  206  is fed to the Cancellation Network  204  where it is subtracted from the delayed output waveform  211  from the main antenna. The resulting difference is an output waveform  207  from the Cancellation Network which is fed back to the ASC and also transmitted to the radar system as is the case in the described application. 
         [0099]    When the invention is implemented in the frequency domain the above described summation and subtraction could alternatively be done for each individual spectral component q followed by one final Inverse Fourier Transform (IFT) forming the output waveform z. Giving the advantage of less required dynamic range since over-excitation of one or a few spectral components will degrade performance gracefully compared to over-excitation in the time domain resulting in a severe performance drop over the entire bandwidth. In this alternative solution there will, instead of one first summation network, be Q single spectral first summation networks, one for each spectral component from the antenna elements in the auxiliary antenna and Q single spectral Cancellation Networks where each spectral component of the delayed output waveform of the main antenna is subtracted with the corresponding spectral component from the auxiliary antenna summarized in respective single spectral first summation network. An IFT is then performed on the waveforms of the resulting spectral components to produce the output waveform z  207 . The advantage with this solution is that an IFT is not required in the transforming means but an FT has to be made of the delayed output waveform  211  from the main antenna to produce the spectral components fed to the single spectral Cancellation Networks. 
         [0000]    The delayed output waveform  211  and the output waveforms from the transforming means can thus be either in the time domain or in the frequency domain divided in Q spectral components. 
         [0100]    The ASC continuously updates the control parameters contained in the matrix W or w, in order to maximize the signal to noise ratio (S/N) of the output waveform  207  from the Cancellation Network  204 . This is accomplished by continuous recalculation of all control parameters as described above. The advantage with using the invention is that the output waveform  209  from the first summation network  206  can be made valid over the entire bandwidth B thereby enabling cancellation of unwanted disturbances picked up from side lobes of the main antenna over the entire operating bandwidth B of the main antenna. 
         [0101]    The angles θ and φ are defined as illustrated in  FIG. 5 . In a Cartesian coordinate system with X-axis  501 , Y-axis  502  and Z-axis  503  the direction to a point  504  in space is defined by an angle θ,  505 , and an angle φ,  506 . The angle φ is the angle between a line  507  from the origin  508  to the point  504  and the Z-axis. The angle θ is the angle between the vertical projection,  509 , of the line  507  on the X-Y plane and the X-axis. 
         [0102]      FIG. 6  shows an antenna pattern  601  of the main antenna and antenna pattern  602  of an antenna element in the auxiliary antenna as a function of the angle θ on the horizontal axis  603 . The vertical axis  604  represents gain in dB. As can be seen the antenna pattern of the main antenna consists of a main lobe  605  and side lobes  606  to  613 . The antenna pattern of the antenna element is fairly omni-directional and at a much lower level than the main lobe of the main antenna. A jammer waveform is now assumed from a jammer source  615  having an incident angle  616 . The ASC is now calculating the control parameters from the waveform W(ω)=A(ω)·e −j·ω·τ(ω)  or W(ω)=A(ω)·e −j·φ(ω)  according to the principles described above such that the amplitude of the antenna pattern in the antenna element in the auxiliary antenna is reduced to an amplitude level representing point  617  at jammer incident angle  616 . This amplitude level corresponds to the amplitude level of the side lobe  607  at the angle  616 . The control parameters are further calculated such that phase of the main antenna and the antenna element will be equal at point  617  by minimizing the residual power in each spectral component q of the output waveform z. The thus modified antenna pattern of the antenna element is shown with the dashed curve  618 . Both gain and phase are now adjusted in the antenna element of the auxiliary antenna to be equal to the main antenna in the desired cancellation direction. The output waveforms s 1 ′ to s N ′ from the antenna elements are fed to the first summation network  206  where they are summarized to a total auxiliary antenna output waveform  209  which is fed to the Cancellation Network where it is subtracted with the delayed output waveform  211  of the main antenna resulting in a waveform cancellation in the direction  616  in the output waveform  207  corresponding to a resulting main antenna pattern  619 . In the embodiment of  FIG. 6  there is just one antenna element in the auxiliary antenna and thus only one output waveform s 1 ′ which is fed directly to the Cancellation Network as there is no need for a first summation network in this embodiment. In another embodiment of the invention, as explained earlier, the first summation network and the Cancellation network can be substituted with Q single spectral first summation networks and Q single spectral Cancellation networks when a final IFT is made after the summation and cancellation calculations. 
         [0103]    The functions of the first summation network can also be integrated in the CN. In this embodiment each output waveform from the transforming means s 1 ′ to s N ′ is directly fed to the CN. 
         [0104]    As can be seen in  FIG. 6  the main lobe of the main antenna will only be marginally affected as the gain level of the antenna elements in the auxiliary antenna is far below the level of the main lobe. The resulting side lobe levels can be either increased or decreased depending on the phase differences between the antenna element and the main antenna at the angle in question. As mentioned earlier the advantage with the invention is that by using the control parameters described above it will be possible to achieve the cancellation over the entire wideband B. 
         [0105]    The method of various embodiments of the invention are shown in  FIG. 7 . All calculations in the embodiment of  FIG. 7  are performed in the ASC. In  701 , after the START the row vector S M  consisting of the elements S Mq  (qε[ 0  . . . Q−1]) being the FT of the output waveform of the main antenna is calculated. In  702  a delay compensation is applied to the row vector S M  by adjusting the phase of each spectral component. In  703 , n is set to 1 and in  704 , row n of the S matrix is calculated. The S matrix consists of the elements S n,q (nε[1 . . . N], qε[0 . . . Q−1]) where S n,q  is the spectral component q of the auxiliary antenna n. In  705 , a check is made if n=N, if not a return is made to  703  where n is increased with 1 and a new row in the S matrix is calculated in  704 . This loop  703  to  705  continues until the check in  705  results in n=N. When this occurs all rows in the S matrix are calculated and the calculations proceed to  706 , where the W matrix consisting of the elements W n,q  (nε[1 . . . N], qε[0 . . . Q−1]) is calculated while minimizing the output residual power P=E[|Z| 2 ]=E[|S M   T −W T ·S| 2 ]. When the W matrix is calculated a new n-loop is started in  707  setting n=1. If a time domain realization is chosen a decision is made in  708  to proceed to  709  where an IFT is made of row n of the W matrix resulting in an output from 709 being the control signal c n . If a time domain realization is not chosen in  708 , i.e. a frequency domain solution is chosen, row n of the W matrix is used in  710  as the control signal c n . When a c n  signal has been produced in either  709  or  710  a check is made in  711  if n=N, if not the loop continues in  707  by increasing the value of n and proceeding in the loop  707 ,  708 ,  709 / 710  and  711  until the check in  711  results in n=N. When n=N all control signals c 1  to c N  have been calculated and the calculation starts all over again by making a new calculation of the row vector S M  in  701 . As explained above the control parameters equal to row  1  to row N of W or row  1  to row N of w depending on the chosen realization of the transforming means, are sent to corresponding transforming means Tr 1  to Tr N  via the control signals c 1  to c N . 
         [0106]    The invention is not limited to the embodiments above, but may vary freely within the scope of the appended claims.