Abstract:
A multiple data-rate receiver uses a signal rate detection technique that employs comparators to obtain information on the incoming data rate for enabling the appropriate receiver. Once a data rate is determined, only then is the appropriate receiver activated. Hence, power dissipation is kept to a minimum during the autonegotiation phase. This is a significant improvement over existing art which require two (or more) receivers to be active during the autonegotiation phase, consequently demanding high power dissipation. Because the autonegotiation phase can be lengthy, the present technique is preferable in many cases.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to a receiver that automatically determines the data rate. 
     2. Description of Prior Art 
     To transmit digital data over a link, such as a twisted-pair copper cable, a transceiver is required. A transceiver contains a local transmitter which broadcasts the digital information over the link and a local receiver which receives information on the link sent by a remote transmitter. The receiver re-creates the information sent by the remote transmitter by compensating for the losses induced by the link on the transmitted signal. Proper operation requires that both transmitter and receiver are configured to transmit and receive information according to a prescribed line code and data rate. For example, in Ethernet applications common transceivers can support 10 Mb/s (Manchester code) or 100 Mb/s (MLT3 code). In multi-rate transceiver circuits, it is possible to support more than one data rate (line code). For example, a “10/100” Ethernet transceiver must accommodate both 10 Mb/s and 100 Mb/s data traffic. Hence, a transceiver must have the capability to decide which mode of operation it is to support for a given link. The decision process is termed “autonegotiation”. During autonegotiation, the two transceivers on opposite ends of the link communicate by transmitting information encapsulated within a periodic burst of closely spaced pulses. These pulses are termed “fast link pulses” (FLPs), and are defined by IEEE standard 802.3. Based on the information conveyed by the pulses (frequency, number), both receivers can decipher what line code capabilities the remote transceiver can support. Once this is known, both transceivers are placed in a line code mode that both can support. If they can support more than one line code, the common capability that is chosen depends on an established priority. 
     Since autonegotiation is a relatively new feature, native transceivers do not support this protocol. Thus another mechanism is required to determine the capability of the link partners. This mechanism involves a signal detect circuit that monitors the incoming signals to decide which line code is being transmitted. Typically if a link is not established, a transceiver is in the idle state, in which case it sends certain distinct patterns. These patterns can be detected by the receiver to decide the remote transmitter line code. Notice that the more line codes one must support, the harder it is to establish which line code is being received based on the data being received. Hence the need for the autonegotiation protocol. The combination of the signal detect mechanism and the autonegotiation mechanism is termed “parallel detection”. 
     When a transceiver is powered up it must constantly remain in this parallel detection mode (i.e. “listen to the wire”). In many situations this state can be lengthy, thus it is necessary to keep power dissipation low during this state as the transceiver is not performing any data transfers but idle. Typical solutions for parallel detection circuitry in 10/100 transceivers is to turn on both the 10 Mb/s receiver and the 100 Mb/s receiver and based on their outputs decide on a common capability between the link partners. Unfortunately, this solution requires significant power. In fact, this idle state dissipates more power than the operating mode, since once a common capability is established, only one receiver is powered up (10 or a 100, not both). 
     FIG. 1 shows an embodiment of a typical 10/100 receive architecture to achieve parallel detection, which is usually implemented on a single integrated circuit (IC) in present-day designs. As used herein, a 10 Base-T transceiver is also referred to as “10BT”, a 100 Base-TX transceiver as “100TX”, and a 100 Base-T4 transceiver as “100BT4”. During autonegotiation both receivers  101  and  102  are active, and supply their output signals to the parallel detect block  103 . The output of the 10BT (10 Mb/s) receiver  101  goes to the FLP block  104 , which monitors the received signal. If proper pulse bursts are detected with the correct frequency and number, the FLP block will determine which of links 100TX or 10BT to establish by asserting or de-asserting the FLP signal, which causes the control block ( 107 ) to assert either the 100TX or the 10BT signal, respectively. An asserted 100TX signal (e.g., high voltage) causes the OR gate  108  to place a “disable” signal on the disable input of the 10BT receiver  101 , thereby allowing only the 100TX receiver  102  to remain active. Alternatively, an asserted 10BT signal causes the OR gate  109  to place a “disable” signal on the disable input of the 100TX receiver  102 , thereby allowing only the 10BT receiver  101  to remain active. If the received signal does not contain proper FLP bursts, a FAIL condition will be reported by the control block  107 . In this case, link information will be obtained from the signal detect blocks; SD ( 105 ) for the 100 Mb/s mode and NLP ( 106 ) for the 10 Mb/s mode. 
     For example, a typical prior-art solution for implementing the SD block employs a peak detector on the post-equalized receiver output. If the post-equalized signal exceeds a certain threshold level for a given time period, the SD output is asserted, indicating a 100 Mb/s link. De-assertion of the SD output occurs if the peak level of the post-equalized output falls below the threshold level in a given time period. Another prior-art technique asserts the SD output when the 100M/bs receiver phase locked-loop (used for timing recovery) acquires lock. Still another prior-art technique makes use of the adaptation output behavior to assert the SD output. In any of these cases, the asserted SD output causes the OR gate  108  to place a disable signal on the disable input of the 10BT receiver section  101 . To check for 10 Mb/s activity, a typical prior-art technique is to check for the idle pattern that this line code provides. These idle patterns are termed “normal link pulses” (NLPs), and are essentially pulses with a duration of 100 ns and with a predictable period, as defined by ISO/IEC 8802-3 and ANSI/IEEE 802.3. If these pulses are detected in a given time period the NLP block  106  asserts the NLP signal. This assertion causes the OR gate  109  to place a disable signal on the disable input of the 100TX receiver section  102 . In the foregoing manner, either the 100TX or 10BT signal indicators will be active so that the transceiver can decipher the capability of its link partner if the autonegotiation protocol fails. Note that an “other” output may be provided from block  107 , to indicate the presence of another signal protocol (e.g., 100BT4). It is evident that the prior-art techniques requires both the 10 Mb/s and the 100 Mb/s receivers to be active during this autonegotiation phase, which can be lengthy as discussed above; thus, power dissipation in this state can be quite high. 
     Referring to FIG. 2, a typical implementation of an analog 100TX receiver shows various components that tend to consume significant amounts of power when activated. The baseline wander block ( 20 ) provides for correction of DC offset voltages due to asymmetrical data streams, while the automatic gain control block ( 21 ) compensates for amplitude differences. These amplitude differences may be due to the insertion loss and variations in the transmitter output, for example. The equalizer ( 22 ) corrects for varying cable lengths and differences in phase delays, and the comparators ( 23 ) provide for detection of the data, while the phase-locked loop ( 24 ) provides for recovery of the clock from the received data. Referring to FIG. 3, a typical implementation of a digital receiver includes an automatic gain control ( 30 ), an equalizer ( 31 ), an analog-to-digital converter ( 32 ) providing digital signal to a digital signal processor ( 33 ) over a bus ( 34 ). While various receiver designs differ in which of these blocks to include, they typically include most of them. 
     SUMMARY OF THE INVENTION 
     We have invented a signal rate detection technique for use in a multi-data rate receiver that uses comparators to obtain information on the incoming data for enabling the appropriate receiver section. A first comparator detects a relatively low data rate signal, and a second comparator detects a relatively high data rate signal. Logic circuitry activates a first receiver section when the first comparator detects the relatively low data rate signal, and activates the second receiver section when the second comparator detects the relatively high data rate signal. 
    
    
     BRIEF DESCRIPTION OF THE DRAWING 
     FIG. 1 shows a typical prior-art multiple data-rate receiver. 
     FIG. 2 shows a typical prior-art analog receiver block diagram. 
     FIG. 3 shows a typical prior-art digital receiver block diagram. 
     FIG. 4 shows an illustrative embodiment of the inventive multiple data-rate receiver. 
     FIG. 5 shows an illustrative signal detection block suitable for use with the inventive receiver. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     The following detailed description relates to a multiple data-rate receiver that determines the rate of incoming data using comparators and logic, thereby avoiding the necessity of maintaining multiple receivers active during the autonegotiation phase. Hence, power dissipation may be kept to a minimum. FIG. 4 shows an illustrative embodiment of the inventive technique for a dual-data rate receiver employing a relatively low data rate receiver section ( 401 ) and a relatively high data rate receiver section ( 402 ). The parallel detection circuitry ( 403 ) comprises two comparators set up with different threshold levels, CMP 10  ( 404 ) for 10 Mb/s data, and CMP 100  ( 405 ) for 100 Mb/s data. CMP 10  is also configured to reject incoming signals that exceed a certain frequency (e.g., 40 MHz). This helps ensure that CMP 10  reacts only to low frequency signals, which are typically the autonegotiation pulse bursts (FLPs), or alternatively the 10BT idle patterns (NLPs). The comparator CMP 10  sets a flip-flop FF 10  ( 406 ) whenever the comparator output is asserted. The flip-flop FF 10  is polled by the controller block ( 407 ) at regular intervals. During the polling period, the FF 10  output is sent to the FLP block ( 408 ) and the NLP block ( 409 ). Once a poll is complete, the controller resets the flip-flop FF 10 . The FLP block makes use of the FF 10  polled outcome to determine if the input sequence meets the criteria (frequency, number) of the FLP pulse bursts. If this is true, then the link is established based on the information in the pulse bursts. If this is not the case, a FAIL condition is indicated, and the transceiver makes use of the NLP or the SD outputs to determine the mode of operation. Once a mode of operation is known, the respective receiver is powered up (enabled). The NLP block works similarly to the FLP block. It makes use of the FF 10  polled outcome to detect if the link integrity pattern is being transmitted, based on periodicity of the idle (NLP) pulses. If so, the 10BT signal is asserted; otherwise, it remains de-asserted. The SD block ( 410 ) determines 100 MB/s activity, which may be either a normal data signal, or the 100 Mb/s idle pattern. If either the NLP signal from block  409  or the 10BT signal from block  407  are active, then the OR gate  411  enables the 10BT receiver ( 401 ). If either the SD signal from block  410  or the 100TX signal from block  407  are active, then the OR gate  412  enables the 100TX receiver ( 402 ). 
     In implementing the above architecture, prior-art circuitry may be used for various of the blocks. For example, the comparators  404  and  405  may be implemented using the technique described in U.S. Pat. No. 5,448,200 co-assigned herewith, with other designs being possible. The flip-flop  406  is of conventional design, and the NLP block  409  may be comparable to that in prior-art FIG.  1 . The FLP block  408  is typically a state machine of conventional design, and the controller  407  is typically dedicated logic of conventional design. The receivers  401  and  402  may be conventional. The SD block  410  may be of a relatively simple design as compared to the prior-art SD block  105  in FIG.  1 . 
     Referring to FIG. 5, an exemplary embodiment of SD block ( 410 ), which determines 100 Mb/s activity, works as follows: The comparator CMP 100  ( 405 ), is used to detect signals exceeding a certain threshold level, VTH. If this takes place, a flip-flop FF 100  ( 501 ) is set. The flip-flop output is polled periodically by the TIMER block ( 502 ). If the flip-flop is set, the assert counter ( 503 ) and de-assert counter ( 504 ) each increment by one count. The flip-flop is reset after the polling phase is complete. The SD output is asserted if the assert counter exceeds a certain count value, CNT1, within a given time period (window), T1. De-assertion of the SD output takes place if the de-assert counter does not exceed a second count value, CNT2, in the time window T2 (note that it is possible to have T1=T2). When the time window expires the respective counters are reset. The parameters VTH, T1, T2, CNT1 and CNT2 are chosen such that given the statistical nature of the 100 Mb/s signal and the attenuation of the cables supported, the SD output will assert/de-assert appropriately. In addition, the parameters T1 and CNT1 are chosen such that 100TX will not assert on 10BT idle patterns or the FLP pulse bursts. This is achieved based on the known frequency, pulse width and number of these pulses. It will be evident to one skilled in the art that both counters can be implemented with a single counter. Furthermore, the above embodiment has shown the comparators  404  and  405  implemented in separate circuit blocks, allowing for independent optimization of the design of each. However, it is alternatively possible to implement the comparison function with the same circuitry; i.e., in a common circuit block. In that case, the following logic circuitry (e.g.,  406  and  410 ) may then receive the comparator output of the common block. 
     While the above embodiment of the invention has been given in terms of a dual-data rate receiver, extension to more than two data rates is possible. For example, the upcoming generation of gigabit (i.e., 1000 MB/s) transceivers may advantageously use the present technique for selectively enabling one of three receivers (i.e., 10, 100, or 1000 MB/s). The present technique may also be combined with other data rate negotiation techniques. For example, a first receiver section may provide for a 10 MB/s data rate, whereas a second receiver section may provide for both 100 MB/s and a 1000 MB/s data rates, which rate may be determined during the autonegotiation process. Still other uses of the inventive technique will be apparent to persons of skill in the art.