Abstract:
An automatic tuning scheme for two active band-pass filters where both filters operate on the signal while simultaneously being tuned using a reference signal. To allow that the amplitude of a reference signal is made small and since both filters demonstrate a good linearity the build-up of the inter-modulation distortion does not occur. The first band-pass filter is tuned with the reference falling into its pass-band. The second band-pass filter is also tuned with the reference placed into its pass-band. The reference is practically eliminated by the virtue of the complexity of the second band-pass filter. Assuming the filter passes the signal for positive frequencies if the reference is made a negative frequency by appropriate 90 degrees phase shifting it will be attenuated by at least 55 dB, which is a sufficient signal-to-reference ratio.

Description:
RELATED APPLICATION 
   This application is a continuation-in-part of U.S. application Ser. No. 10/423,923 filed Apr. 28, 2003 and entitled “Fully Integrated Automatically Tuned FR and IF Active Bandpass Filters” and claims the benefit therefrom. 

   TECHNICAL FIELD OF THE INVENTION 
   The present invention relates to fully-integrated continuous-time active band-pass filters and their automatic frequency- and Q-tuning systems. 
   BACKGROUND OF THE INVENTION 
   Nearly all practical transceivers require some form of filtering. Up to the date, in majority of cases, these radio-frequency (RF) and intermediate-frequency (IF) filters are realized off-chip as ceramic or surface-acoustic wave (SAW) devices. 
   A first reason for slow progress in integration RF and IF filters is their rather modest noise and distortion performance. This can be alleviated by a careful overall system design by taking into account the filter short comings and by purposely reducing their requirements while simultaneously offsetting their reduced performance with superior performance of preceding and following high-quality blocks. 
   A second reason for slow progress in integration RF and IF filters is that these filters require circuitry for adjusting their center or corner frequency as well as their quality-, or (Q)-factors. The accuracy problems of such tuning systems may result in the whole filter not meeting the stringent system specifications over process, voltage supply and temperature variation (PVT). Due to the matching errors the most frequently used Master-Salve (M-S) schemes suffer from significant accuracy errors averaging up to 5% for frequency schemes and up to several tens of percent for Q-tuning schemes. Naturally, for majority radio applications such a modest accuracy is not acceptable. Additionally, the reference feed-through degrades the overall noise performance of the filter. The typically achievable S/N ratio of an active filter tuned with a M-S scheme is about 40 dB. 
   A third reason for slow progress in integration RF and IF filters is taking full advantage of possible system and circuit trade-offs during the transceiver design. In order to make a design of RF or IF filters viable, their system specification should be optimized and carefully negotiated with the overall system specifications. In other words, for a successful implementation of a fully-integrated transceiver the sequence of specification building should be reversed: knowing the limitations of the active filters one should design the system architecture, its system specifications and other circuits to alleviate these shortcomings. Only then the whole system has a chance to meet its overall requirements. 
   In the case of the presented RF and IF filters the tuning accuracy can be substantially improved with tuning the filter signal directly instead of using the Master-Slave (M-S) scheme by passing the reference through it while simultaneously processing the signal. The expected accuracy of such frequency- and Q-tuning systems could reach 0.5% and 2% respectively. There are certain requirements for the reference signal that need to be fulfilled:
         its frequency should fall at the edge of the pass-band of the RF filter, but its frequency should be chosen so that it will not inter-modulate with the adjacent channel carrier;   the reference amplitude of the FR filter should be at least 15 dB lower than the selected channel carrier;   given these conditions the reference passes the RF band-pass (BP) filter linearly without causing inter-modulation distortion. Suppose that the reference frequency it is Δf away from the desired channel carrier. Give the attenuation characteristic of the IF BP filter Δf separation should be chosen such that the IF BP filter attenuates the reference of the RF BP filter by at least 40 dB;   hence, at the output the signal and the reference differ at least by 55 dB, which is better than the reference feed-through of a typical M-S scheme;   the reference of the IF BP filter is rejected by its appropriate conditioning with respect to the main signal and the complex nature of the IF BP filer. The expected attenuation of the IF BP filter reference is at least 55 dB.       

   DESCRIPTION OF THE PRIOR ART 
   The architecture of a classical Master-Slave (M-S) tuning scheme such as one described in U.S. Pat. No. 3,997,856 is illustrated in  FIG. 1 , and is identified by the numeral  10 . Note that only a frequency-tuning scheme is presented in U.S. Pat. No. 3,997,856. The Q-tuning scheme is not disclosed in that patent. The frequency-control part consists of a Master filter (or Master oscillator)  12  followed by the frequency-tuning circuit  14  that similarly to U.S. Pat. No. 3,997,856 may consist of a phase-detector, a low-pass filter and a differential amplifier. These blocks are not drawn separately here for the sake of simplicity. The Q-control part of the scheme consists of the same Master filter (or Master oscillator)  12  followed by the Q-tuning circuit  16 . It may consist of an amplitude detection circuit and a differential amplifier. Again, for the sake of simplicity these blocks are not drawn separately here. Since the frequency- and Q-control loops directly control the Master circuit  12  and not the Slave filter  18  due to unavoidable on-chip matching errors the expected accuracy of the M-S frequency scheme could be as low as 5% for frequency-tuning and up to several tens of percent for Q-tuning. 
   The architecture of a typical filter wafer-probe trimming scheme is illustrated in  FIG. 2 , and is identified by the numeral  20 . It consists of the g m -C oscillator circuit  22  with its output connected to the phase detector  24 . The frequency reference is applied to the second input of the phase detector  24 . The difference signal at the output of the phase detector  24  is low-pass filtered by filter  26  and applied to the input of a high-gain amplifier that controls g m  of the oscillator circuit  22 . During the one-time wafer-probe procedure the oscillation frequency is measured and compared to the frequency reference. The resulting error is used by a negative feedback loop to adjust the g m  of the oscillator, which is built with the same g m -amplifiers as the tuned filter. After the filter g m  amplifiers are adjusted their value becomes fixed, the oscillator and the rest of the trimming circuitry are disabled and the filter g m  amplifiers and its frequency track the temperature by the virtue of its temperature independent biasing. The accuracy of the wafer-probe scheme is expected to be worse than the M-S scheme. The reason is that the wafer-probe scheme demonstrates the M-S accuracy at the beginning, but then after the Master is disabled it relies entirely on the biasing. Since the biasing introduces an extra error by not being able to track the temperature variation perfectly the overall accuracy of the wafer-probe scheme can be as low as 10% for frequency-tuning, which is still useful for some less demanding applications. However, because of the low accuracy, the Q-tuning is not practical using wafer-probe scheme. 
   The architecture of a proposed self-tuned filter scheme is illustrated in  FIG. 3 , and is identified by the numeral  30 . It consists of two filters  31  and  32 . Using switches  33  and  34 , when the first of filters  31  and  32  has its input connected to the input signal the other one is tuned and then their roles are interchanged. The outputs of filters  31  and  32  are switched using switches  35  and  36 . The frequency reference is applied to the frequency-tuning circuit  39  that generates control signals via hold circuits  37  and  38  for tuning the one of the filters  31  and  32  that is not processing the input signal. The critical difficulty of this scheme is switching the filters on and off the signal such that transients are avoided. As far as accuracy is concerned the proposed self-tuning achieves higher-accuracy than that of M-S scheme. Its tuning accuracy error is related to the overall error of the frequency-tuning scheme, which can be as low as 0.5%. 
   SUMMARY OF THE INVENTION 
   The present automatic tuning scheme for two BP filters is used in a fully-integrated heterodyne receiver such as one in  FIG. 16 . Contrary to other tuning schemes both filters operate on the signal while simultaneously being tuned using a reference signal. To allow that the amplitude of reference signal is made small and since both filters demonstrate a good linearity the build-up of the inter-modulation distortion does not occur. For accuracy reasons the first BP is tuned with the reference falling into its pass-band. Under such conditions the reference is not practically attenuated by the first BP filter. However the same distance between the pass-band center and the reference frequency in the second BP filter results in about 40 dB attenuation of the first BP reference. This is because the second BP bandwidth is much narrower than the first BP bandwidth. At lower IF frequency its order and its Q&#39;s can be made sufficiently high to achieve sufficient attenuation. This combined with at least 15 dB the original signal-to-reference ratio results in the desired 55 dB the final signal-to-reference ratio. However, the second band-pass is also tuned. Since there is no further filter in the signal path its tuning reference cannot be removed. To solve this problem the complexity of the second BP is exploited. Assuming the filter passes the signal for positive frequencies if the reference is made a negative frequency by appropriate 90 degrees phase shifting it will be attenuated by at least 55 dB, which is a sufficient signal-to-reference ratio. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     For a more complete understanding of the present invention and for further advantages thereof, reference is now made to the following Description of the Preferred Embodiments taken in conjunction with the accompanying Drawings in which: 
       FIGS. 1 ,  2  and  3  are block diagrams of prior art tuning schemes; 
       FIG. 4  is a block diagram of the present automatically tuned band-pass filter equipped with frequency- and Q-tuning schemes that operate while the filter is processing the signal; 
       FIG. 5  illustrates the concept of a tuning scheme for the real RF BP filter; 
       FIG. 6  illustrates the concept of tuning scheme for the complex IF BP filer; 
       FIG. 7  presents more detailed arrangement for the tuning scheme of the real RF BP filter built with second-order biquad sections; 
       FIG. 8  presents more detailed arrangements for the tuning scheme the complex IF BP filter built with second-order biquad sections; 
       FIG. 9  presents the schematic of implementation of frequency-tuning scheme used both in the real RF BP and complex IF BP filters; 
       FIG. 10  presents the schematic of implementation for the systems generating low-frequency reference used in Q-tuning scheme of the complex IF BP filter; 
       FIG. 11  illustrates the concept of using low-frequency reference for the purpose of Q-tuning in the complex IF BP filter; 
       FIG. 12  presents the voltage dividers used to condition the low-frequency reference if the case of sharing such a reference by many biquads in the Q-tuning scheme of the complex IF BP filter; 
       FIG. 13  presents the schematic of implementation of Q-tuning scheme used both in the real RF BP and complex If BP filters; 
       FIGS. 14   a  and  14   b  illustrate the attenuation of the signal and the tuning reference of the first BP filter, after passing first BP filter and second BP filter; 
       FIG. 15   a  illustrates the attenuation of the complex second BP filter for I,Q signal conditioning: the signal is passed for positive frequencies and blocked for negative frequencies (image rejection); 
       FIG. 15   b  illustrates the attenuation of complex nature of the second BP filter for I,-Q signal conditioning; the IF BP filter references is passed for negative frequencies and blocked for positive frequencies; and 
       FIG. 16  is a block diagram of the fully integrated heterodyne receiver using the present automatically tuned BP filters. 
   

   DESCRIPTION OF THE PREFERRED EMBODIMENTS 
   Referring to the  FIG. 4 , the present automatically-tuned filter system is illustrated, and is generally identified by the numeral  40 . The input and reference signal enter the filter circuit  42 , one of the filter outputs is connected to the frequency-tuning block  44  and the other one is connected to the Q-tuning block  46 . The output of the frequency-tuning block  44  is then fed back to the filter  42  in order to control its frequency. Similarly, the output of the Q-tuning block  46  is then fed back to the filter  42  in order to control its Q. 
   Referring to the  FIG. 5 , the present automatically-tuned real RF BP filter system is illustrated, and is generally identified by the numeral  40   a . The input and reference signals enter the filter circuit  42   a  and the reference signal enters the frequency-tuning block  44   a  and the Q-tuning block  46   a . One of the filter outputs is connected to the frequency-tuning block  44   a  and the other is connected to the Q-tuning block  46   a . The output of the frequency-tuning block  44   a  is fed back to the filter  42   a  in order to control its frequency. Similarly, the output of the Q-tuning block  46   a  is fed back to the filter  42   a  in order to control its Q-factor. It should be noted that the filter input, outputs and the reference signals as well as the frequency- and Q-control signals may be single-ended as indicated for its simplicity in  FIG. 5 , but in most practical cases they will be differential, or balanced around a dc common-mode voltage. 
   Referring to the  FIG. 6 , the present automatically-tuned complex IF BP filter system is illustrated, and is generally identified by the numeral  40   b . The input I, Q signals and the reference I, -Q signals enter the filter circuit  42   b . The reference I signal enters the frequency-tuning block  44   b  and the low-frequency reference I signal enters the Q-tuning block  46   b . One of the filter I outputs is connected to the frequency-tuning block  44   b  and the other I output is connected to the Q-tuning block  46   b . The output of the frequency-tuning block  44   b  is fed back to the filter  42   b  in order to control its frequency. Similarly, the output of the Q-tuning block  46   b  is fed back to the filter  42   b  in order to control its Q-factor. It should be noted that the filter I, Q inputs, outputs and the Q, I reference signals as well as the frequency- and Q-control signals may be single-ended as indicated for its simplicity in  FIG. 6 , but in most practical cases they will be differential, or balanced around a dc common-mode voltage. 
   Referring to the  FIG. 7 , the present automatically-tuned real RF BP filter system is illustrated, and is generally identified by the numeral  40   c . The filter is built as a cascade of N second-order sections (biquads). Since each biquad have a different center frequency, for the accuracy purpose they need to be tuned separately. Therefore, each of the filter biquads  411 ,  412  has its own dedicated frequency-  431 ,  432  and Q-tuning  451 ,  452  schemes as well as a separate reference signal. The input and reference signals enter each of the filter biquads  411 ,  412  and the reference signals enter each the frequency-tuning blocks  431 ,  432  and the Q-tuning block  451 ,  452 . The LP output of each biquad is connected to the frequency-tuning block  431  or  432  and the BP output is connected to the Q-tuning block  451  or  452 . The output of the frequency-tuning blocks  431  and  432  is fed back to the appropriate biquad  411  or  412  in order to control its frequency. Similarly, the output of the Q-tuning blocks  451  or  452  is fed back to the appropriate biquad  411  or  412  in order to control its Q-factor. It should be noted that the filter input, outputs and the reference signals as well as the frequency- and Q-control signals may be single-ended as indicated for its simplicity in  FIG. 7 , but in most practical cases they will be differential, or balanced around a dc common-mode voltage. 
   Referring to the  FIG. 8 , the present automatically-tuned complex IF BP filter system is illustrated, and is generally identified by the numeral  40   d . The filter is built as a cascade of N complex second-order sections (biquads). Since each biquad have a different center frequency, for the accuracy purpose they need to be tuned separately. Therefore, each of the filter biquads  421 ,  422  has its own dedicated frequency-  441 ,  442  and Q-tuning  461 ,  462  schemes as well as a separate reference signal. The input I, Q signals and reference I, -Q signals enter each of the filter biquads  421 ,  422 , the reference I signals enter each of the frequency-tuning blocks  441 ,  442  and the low-frequency reference I signals enter each of the Q-tuning blocks  461 ,  462 . The I LP output of each biquad is connected to the frequency-tuning block  441  or  442  and the I BP output is connected to the Q-tuning block  461  or  462 . The output of the frequency-tuning block  441  or  442  is fed back to the appropriate biquad  421  or  422  in order to control its frequency. Similarly, the output of the Q-tuning block  461  or  462  is fed back to the appropriate biquad  421  or  422  in order to control its Q-factor. It should be noted that the filter I, Q inputs, outputs and the I, -Q reference signals as well as the frequency- and Q-control signals may be single-ended as indicated for its simplicity in  FIG. 8 , but in most practical cases they will be differential, or balanced around a dc common-mode voltage. 
   The output of the filter  42  may also serve as a frequency- and/or Q-tuning output. In such a case the original frequency- or Q-tuning output(s) is (are) redundant and is (are) not used. 
   Referring to the  FIG. 9 , the frequency-tuning for an automatically-tuned real RF BP or a complex IF BP filter systems is illustrated, and is generally identified by the numeral  40   e . A pair of limiters  445  limits the frequency-output signal and the reference signal. Their outputs are connected to a phase-detector  446  that can be a sequential detector or a XOR/XNOR gate. The output of the phase detector  446  is filtered by a lowpass filter  447 . The output of filter  447  (Frequency control) is fed back to the filter  42   a  ( FIG. 5) and 42   b  ( FIG. 6 ) in order to control the frequency of filters  42   a  and  42   b . The frequency-tuning block in  FIG. 9  generates a voltage that is proportional to the frequency of the filter. As an example in g m -C filters, the filter frequency is proportional to the ratio g m /C of transconductance g m  and the capacitance C. By changing the voltage that controls the bias of transconductors and their g m -value the frequency tuning effect is achieved. Other standard techniques to tune other types of analog filters using a voltage can be also used. It should be noted that the filter output, the reference signals and the frequency-control signals may be single-ended as indicated for its simplicity in  FIG. 9 , but in most practical cases they will be differential, or balanced around a dc common-mode voltage. In such a case the limiters  445 , the phase-detector  446  and the lowpass filter  447  need to be realized as fully differential or balanced blocks. 
   Referring to the  FIG. 10 , the low-frequency reference generating circuit for an automatically-tuned complex IF BP filter Q-tuning system is illustrated, and is generally identified by the numeral  40   f . It consists of a low-frequency complex biquad  471  and a frequency-tuning  481 . The frequency of the low-frequency reference is chosen to be so low that the excess phase of biquad  471  is negligible and its effective Q is close to the designed Q. Also, since the low-frequency reference biquad is a replica of the filter biquad their image suppression will be similar. The only difference between the attenuation of the low and high reference frequencies is the Q-enhancement of the measured filter. 
   Referring to the  FIG. 11 , the transfer functions for a low- and high-frequency complex biquads for I, Q signals and I, -Q references (image) are illustrated. The main biquad operates at much higher frequency than the low-frequency biquad. Its effective Q*=Q/(1−2*Q*Δφ)=Q/(1−2*Q*(ω o /ω p )), where Q is the designed Q, Δφ is its excess phase, ω ch  is the its pole frequency and ω p  is the pole frequency of its transconductors. Since at the low-frequency the excess phase Δφ=0 the effective Q of the low-frequency biquad is Q*≈Q. The negative feedback of the Q-tuning system brings the difference between the magnitudes of the biquad image transfer functions ΔQ imag  to zero. It is equivalent to compensate for the excess phase errors of the main biquad. 
   Referring to the  FIG. 12 , the reference voltage-dividers for an automatically-tuned complex IF BP Q-tuning system are illustrated, and are generally identified by the numeral  40   h . Depending on the required accuracy and available power budget the system may contain one or more low-frequency biquads. Ideally, each filter biquad should have its own low-frequency reference biquad. In such a case the dividers  40   h  are not required. However, if low-frequency biquad is shared by two or more biquads the dividers are necessary. If the Q of the given biquad is Q 2  and the Q of the low-signal biquad is Q 1  then the resistor ratios can be easily calculated from Q 2 /Q 1 =(R 2 /(R 1 +R 2 ))/(R 3 /(R 1 +R 3 ))=(R 2 /R 3 )*(1+(R 1 /R 2 ))/(1+(R/R 3 )). The buffers  495  are added to eliminate the loading effects. 
   Referring to the  FIG. 13 , the Q-tuning for automatically-tuned real RF BP and complex IF BP filter systems is illustrated, and is generally identified by the numeral  40   i . The Q-output signal and the low-frequency reference signal are peak-detected by a pair of peak-detectors  465 . Their outputs are connected to a differential amplifier  466 . The output of the differential amplifier is filtered by a lowpass filter  467  and fed back to the filter in order to control its Q-factor. It should be noted that the filter output, the low-frequency reference signals and the Q-control signals may be single-ended as indicated for its simplicity in  FIG. 13 , but in most practical cases they will be differential, or balanced around a dc common-mode voltage. In such a case the peak-detectors  465 , the differential amplifier  466  and the lowpass filter  467  need to be realized as fully differential or balanced blocks. 
   If the filter circuit  42  is a real band-pass filter such as filter  74  illustrated in  FIG. 16  the reference is placed at frequency f 2  that is offset from the filter center frequency f 1  by Δf=f 2 −f 1 . This situation is illustrated in  FIG. 14   a . Since the reference passes close to the pass-band its amplitude is kept low to avoid any inter-modulation distortion. The reference is attenuated only by about 1–2 dB by filter  74 . However, as shown in  FIG. 14   b  filter  77  ( FIG. 16 ) attenuates the reference by at least 40 dB. This is due to the fact that filter  77  is much narrower than filter  74  and since Δf=f 4 −f 3  the reference falls into the filter stop-band. Additionally, since the center frequency of filter  77  is much lower than the of filter  74 , f 3 &lt;&lt;f 1 , it is easier to implement higher Q&#39;s and steeper roll-off for filter  77 . 
   If the filter circuit  42  is a complex BP filter, depending on the input signal conditioning, filter  42  passes certain signals and suppresses the others. Assuming the main signal is passed in the form of I and Q components, with Q lagging I by 90 degrees, if the reference is passed in the form of I and -Q components, with Q leading I by 90 degrees, then the filter  42  attenuates the reference by at least 55 dB compared to the main signal. 
   As illustrated in  FIG. 15   a  the main signal characterized by I and Q components is passed by the filter  77  ( FIG. 16 ) for positive frequencies. However, the reference characterized by I and -Q components is suppressed by the filter  77 . Naturally, since the reference is I,-Q conditioned, as presented in  FIG. 15   b , it passes the filter for the negative frequencies, but it is blocked for the positive frequencies such that it does not appear with the main signal. 
   The filter is directly tuned with reference signal while simultaneously operating on the main signal. By choosing appropriate input amplitude of the reference, the reference output amplitude is set to be sufficiently small to not interfere with the main signal for a given type of signal modulation. 
   Any viable frequency-tuning technique including, but not limited to phase detection used in phase locked-loop Type I, or phase and frequency detection used in phase locked-loop Type II can be used to implement frequency-tuning circuit  44 . 
   Any viable Q-tuning technique including, but not limited to amplitude detection using rectifiers and envelope-detectors can be used to implement frequency-tuning circuit  46 . 
   The phase, phase/frequency, delay, or amplitude-locked loops used in frequency- and Q-tuning circuits  44  and  46  can be analog, mixed-mode, digital or software. 
     FIG. 16  illustrates a fully integrated heterodyne receiver  70  using the present filters  74  and  77 . The signal from the antenna enters the input of the low-noise amplifier (LNA) circuit  72 , the output of which is connected to filter  74 . The output of filter  74  is then connected to the two inputs of the complex mixer circuit  76 , which consists of two identical mixers fed by identical input signals and two LO signals shifted by 90 degrees (LO I and LO Q). The complex mixer has two outputs I and Q. They enter two inputs of filter  77 . The I and Q outputs of filter  77  are connected to the input of the variable gain amplifier (VGA) circuit  78 .