Abstract:
A directive antenna and method of directing a radio frequency wave received by and/or transmitted from the antenna. The antenna preferably includes a high impedance surface with a plurality of antenna elements disposed on said surface, a plurality of associated demodulators and power sensors and a switch. A Vivaldi Cloverleaf antenna is disclosed.

Description:
TECHNICAL FIELD 
     The present invention relates to a new antenna apparatus. The antenna apparatus is directional and the receiving and transmitting portion thereof preferably of a thin, flat construction. The antenna has multiple elements which provide directivity. The antenna may be flush-mounted on a high impedance surface. The antenna apparatus includes beam diversity hardware to improve the signal transmission and reception of wireless communications. Since the receiving/transmitting portion of the antenna apparatus antenna may be flush-mounted, it can advantageously used on a mobile platform such as an automobile, a truck, a ship, a train or an aircraft. 
     BACKGROUND OF THE INVENTION 
     Prior art antennas and technology includes: 
     T. Schwengler, P. Perini, “Combined Space and Polarization Diversity Antennas”, U.S. Pat. No. 5,923,303, Jul. 13, 1999. An antenna system with both spatial and polarization diversity has a first antenna aperture and a second antenna aperture, with a polarization separation angle being formed by the difference between the polarization angle of the first antenna aperture and the polarization angle of the second antenna aperture, and a vertical separation being formed by mounting the second antenna aperture a vertical distance above the first antenna aperture, such that diversity gain is achieved by both the polarization angle and the vertical distance. The combination of spatial and polarization diversity allows closer antenna aperture spacing and non-orthogonal polarization angles. However, using current techniques, antennas having both polarizations cannot lie in a single plane—so the resulting antenna is not a low-profile antenna like the antenna disclosed herein. 
     M. Schnetzer, “Tapered Notch Antenna Using Coplanar Waveguide” U.S. Pat. No. 5,519,408. Tapered notch antennas, which are sometime known as Vivaldi antennas, may be made using standard printed circuit technologies. 
     D. Sievenpiper, E. Yablonovitch, “Circuit and Method for Eliminating Surface Currents on Metals” U.S. Provisional patent application, Ser. No. 60/079,953, filed on Mar. 30, 1998. 
     It is also known it the prior art to place a conformable end-fire or array on a Hi-Z surface. It has been shown that the Hi-Z material can allow flush-mounted antennas to radiate in end-fire mode, with the radiation exiting the surface at a small angle with respect to the horizon. 
     Conventional vehicular antennas consist of a vertical monopole which protrudes from the metallic exterior of vehicle, or a dipole embedded in the windshield or other window. Both antennas are designed to have an omnidirectional radiation pattern so signals from all directions can be received. One disadvantage of omnidirectional antennas is that they are particularly susceptible to interference and fading, caused by either unwanted signals from sources other than the desired base station, or by signals reflected from vehicle body and other objects in the environment in a phenomenon known as multipath. Antenna diversity, in which several antennas are used with a single receiver, can be used to help overcome multipath problems. The receiver utilizing antenna diversity switches between the antennas to find the strongest signal. In more complicated schemes, the receiver can select a linear combination of the signals from all antennas. 
     The disadvantage of antenna diversity is the need for multiple antennas, which can lead to an unsightly vehicle with poor aerodynamics. Many geometries have been proposed which reduce the profile of the antenna, including patch antennas, planar inverted F-antennas, slot antennas, and others. Patch and slot antennas are described by, C. Balanis,  Antenna Theory, Analysis and Design,  2nd ed., John Wiley &amp; Sons, New York (1997). Planar inverted F-antennas are described by M. A. Jensen and Y. Rahmat-Samii, “Performance analysis of antennas for handheld transceivers using FDTD,”  IEEE Trans. Antennas Propagat.,  vol. 42, pp. 1106-1113, August 1994. These antennas all tend to suffer from unwanted surface wave excitation and the need for thick substrates or cavities. 
     As such, there is a need for an antenna which has low profile and has sufficient directivity to take advantage of antenna diversity. Preferably the antenna should not suffer from the effects of surface waves on the metal exterior of the vehicle. 
     The high impedance (Hi-Z) surface, which is the subject of U.S. No. 60/079,953 mentioned above, provides a means of fabricating very thin antennas, which can be mounted directly adjacent to a conductive surface without being shorted out. Near the resonance frequency, the structure exhibits high electromagnetic impedance. This means that it can accommodate non-zero tangential electric fields at the surface of a low-profile antenna, and can be used as a shielding layer between the metal exterior of a vehicle and the antenna. The total height is typically a small fraction of a wavelength, making this technology particularly attractive for mobile communications, where size and aerodynamics are important. Another property of this Hi-Z material is that it is capable of suppressing the propagation of surface waves. Surface waves normally exist on any metal surface, including the exterior metal skin of a vehicle, and can be a source of interference in many antenna situations. Surrounding the antenna with a small area of Hi-Z surface can shield the antenna from these surface waves. This has been shown to reduce multipath interference caused by scattering from ground plane edges. 
     The present application is related to (i) U.S. patent application Ser. No. 09/537,923 entitled “A Tunable Impedance Surface” filed Mar. 27, 2000, (ii) U.S. patent application Ser. No. 09/537,922 entitled “An Electronically Tunable Reflector” filed Mar. 29, 2000, (iii) U.S. patent application Ser. No. 09/537,921 entitled “An End-Fire Antenna or Array on Surface with Tunable Impedance” filed Mar. 29, 2000, (iv) U.S. patent application Ser. No. 09/520,503 entitled “A Polarization Converting Radio Frequency Reflecting Surface” filed Mar. 8, 2000, and to (v) U.S. patent application Ser. No. 09/525,832 entitled “Vivaldi Cloverleaf Antenna” filed Mar. the disclosures of which are hereby incorporated herein by this reference. 
     The Hi-Z surface, which is the subject matter of U.S. patent application Ser. No. 60/079,953 and which is depicted in FIG. 1 a , includes an array of resonant metal elements  12  arranged above a flat metal ground plane  14 . The size of each element is much less than the operating wavelength. The overall thickness of the structure is also much less than the operating wavelength. The presence of the resonant elements has the effect of changing the boundary condition at the surface, so that it appears as an artificial magnetic conductor, rather than an electric conductor. It has this property over a bandwidth ranging from a few percent to nearly an octave, depending on the thickness of the structure with respect to the operating wavelength. It is somewhat similar to a corrugated metal surface  22  (see FIG. 1 b ), which has been known to use a resonant structure to transform a short circuit into an open circuit. Quarter wavelength slots  24  of a corrugated surface  22  are replaced with lumped circuit elements in the Hi-Z surface, resulting in a much thinner structure, as is shown in FIG. 1 a.  The Hi-Z surface can be made in various forms, including a multi-layer structure with overlapping capacitor plates. Preferably the Hi-Z structure is formed on a printed circuit board (not shown in FIG. 1 a ) with the elements  12  formed on one major surface thereof and the ground plane  14  formed on the other major surface thereof. Capacitive loading allows a frequency be lowered for a given thickness. Operating frequencies ranging from hundreds of megahertz to tens of gigahertz have been demonstrated using a variety of geometries of Hi-Z surfaces. 
     It has been shown that antennas can be placed directly adjacent the Hi-Z surface and will not be shorted out due to the unusual surface impedance. This is based on the fact that the Hi-Z surface allows a non-zero tangential radio frequency electric field, a condition which is not permitted on an ordinary flat conductor. 
     In one aspect the present invention provides an antenna apparatus for receiving and/or transmitting a radio frequency wave, the antenna apparatus comprising: a high impedance surface; an antenna comprising a plurality of flared notch antennas disposed immediately adjacent said surface; a plurality of demodulators with each of said plurality of demodulators being coupled to an associated one of said plurality of flared notch antennas; a plurality of power sensors with each of said plurality of power sensors being coupled to an associated one of said plurality of demodulators; and a power decision circuit responsive to outputs of said power sensors for coupling selected one of said plurality of antennas to an output. 
     In another aspect the present invention provides an antenna apparatus for receiving and/or transmitting a radio frequency wave, the antenna apparatus comprising: a high impedance surface; an antenna comprising a plurality of flared notch antennas disposed immediately adjacent said surface; at least one demodulator coupled to said plurality of flared notch antennas; at least one power sensor coupled to said at least one demodulator; and a power decision circuit responsive to outputs of said at least one power sensor for coupling selected one of said plurality of antennas to an output. 
     In yet another aspect the present invention provides an antenna apparatus for receiving and/or transmitting a radio frequency wave, the antenna comprising: a plurality of flared notch antennas disposed adjacent to each other and arranged such that their directions of maximum gain point in different directions, each of the flared notch antennas being associated with a pair of radio frequency radiating elements and wherein each radio frequency radiating element serves as a radio frequency radiating element for two different flared notch antennas. The apparatus also includes a plurality of demodulators with each of said plurality of demodulators being coupled to an associated one of said plurality of flared notch antennas; a plurality of power sensors with each of said plurality of power sensors being coupled to an associated one of said plurality of demodulators; and a power decision circuit responsive to outputs of said power sensors for coupling selected one of said plurality of antennas to an output. 
     In still yet another aspect the present invention provides a method of receiving and/or transmitting a radio frequency wave at an antenna apparatus comprising: a high impedance surface and an antenna comprising a plurality of antennas disposed immediately adjacent said surface such that, the method comprising the steps of: (a) demodulating signals from said antennas; (d) sensing power of signals from said antennas; and (e) coupling said plurality of antennas to an output as a function of the sensed power of signals from said antennas. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 a  is a perspective view of a Hi-Z surface; 
     FIG. 1 b  is a perspective view of a corrugated surface; 
     FIG. 1 c  is an equivalent circuit for a resonant element on the Hi-Z surface; 
     FIG. 2 is a plan view of a Vivaldi Cloverleaf antenna according to one aspect of the present invention; 
     FIG. 2 a  is a detailed view of the Vivaldi Cloverleaf antenna at one of its feed points; 
     FIG. 3 depicts the Vivaldi Cloverleaf antenna disposed against a Hi-Z surface in plan view; 
     FIG. 4 is a elevation view of the antenna and Hi-Z surface shown in FIG. 3; 
     FIG. 5 is a schematic plan view of a small portion of a three layer high impedance surface; 
     FIG. 6 is a side elevational view of the three layer high impedance surface of FIG. 5; 
     FIG. 7 is a plot of the surface wave transmission magnitude as a function of frequency for a three layer high impedance surface of FIGS. 5 and 6; 
     FIG. 8 is a graph of the reflection phase of the three layer high impedance surface of FIGS. 5 and 6 plotted as a function of frequency; 
     FIG. 9 is a graph of the elevation pattern of a beam radiated from a flared notch of a Vivaldi Cloverleaf antenna disposed on the high impedance surface of FIGS. 5 and 6; 
     FIG. 10 is a graph of the radiation pattern taken through a 30 degree conical azimuth section of the beam transmitted from a flared notch of a Vivaldi Cloverleaf antenna disposed on the high impedance surface of FIGS. 5 and 6; 
     FIG. 11 is a system diagram of the low profile, switched-beam diversity antenna; 
     FIG. 12 depicts the electric fields that are generated by exciting one the flared notch antenna in the upper left hand quadrant of the Vivaldi Cloverleaf antenna; 
     FIG. 13 depicts the radiation pattern when the feed point for the upper left hand quadrant of the Vivaldi Cloverleaf antenna is excited; 
     FIG. 14 depicts the wires antenna elements disposed against a Hi-Z surface in plan view; 
     FIG. 15 is a elevation view of the antenna and Hi-Z surface shown in FIG. 14; 
     FIG. 16 is a graph of the elevation pattern of a beam radiated from a wire antenna disposed on the high impedance surface of FIGS. 5 and 6; 
     FIG. 17 is a graph of the radiation pattern taken through a 30 degree conical azimuth section of the beam transmitted from a flared notch of a wire antenna disposed on the high impedance surface of FIGS.  5  and  6 . 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     The present invention provides an antenna, which is thin and which is capable of switched-beam diversity operation for improved antenna performance in gain and in directivity. The switched-beam antenna design offers a practical way to provide an improved signal/interference ratio for wireless communication systems operating in a mobile environment, for example. The antenna may have a horizontal profile, so it can be easily incorporated into the exterior of vehicle for aerodynamics and style. It can be effective at suppressing multipath interference, and it can also be used for anti-jamming purposes. 
     The antenna includes an array of thin antenna elements, or sub-arrays, which are preferably mounted on a Hi-Z ground plane. The Hi-Z ground plane provides two features: (1) it allows the antenna to lie directly adjacent to the metal exterior of the vehicle without being shorted out and (2) it can suppress surface waves within the operating band of the antenna. 
     The antennas can be arrays of Yagi-Uda antennas, slot antennas, patch antennas, wire antennas, Vivaldi antennas, or preferably, if horizontal polarization is desired, the Vivaldi Cloverleaf antenna disclosed herein. Each individual antenna or group of antenna elements, in the case of Yagi-Uda antennas, preferably have a particular directivity (sometimes corresponding to the number of elements utilized) and this directivity impacts the number of beams which can be conveniently used. For example, the total omnidirectional radiation pattern can be divided into several sectors with different antennas addressing different sectors. Each individual antenna (or group of antenna elements as in the case of Yagi-Uda antennas) in the array can then address a single sector. Thus, a four antennas may be used in an array if each such antenna has a directivity that is four times better than an omnidirectional monopole antenna. 
     FIG. 2 is a plan view of an antenna  50  formed of an array or group of four antenna elements  52 A,  52 B,  52 C and  52 D which in effect form four different antennas. The four elements  52  have four feed points  54 A,  54 B,  54 C and  54 D therebetween and the antenna  50  has four different directions  56 A,  56 B,  56 C and  56 D of greatest gain, one associated with each feed point. However, the antenna may have more than or fewer than four elements  52 , if desired, with a corresponding change in the number of feed points  54 . The impedance at a feed point is compatible with standard 50Ω radio frequency transmitting and receiving equipment. The number of elements  52  making up the antenna is a matter of design choice. While the inventors have only made antennas with four elements  52  to date, they expect that antennas with a greater number of elements  52  could be designed to exhibit greater directivity, but would require a larger area and a greater number of feed points. Those skilled in the art will appreciate that better directivity could be an advantage, but that larger area and a more complex feed structure could be undesirable for certain applications. 
     FIG. 2 a  is a detailed partial view of two adjacent elements  52  and the feed point  54  therebetween. The feed points  54  are located between adjacent elements  52  and conventional unbalanced shielded cable may be used to couple the feed points to radio frequency equipment used with the antenna. 
     Each element  52  is partially bisected by a gap  58 . The gap  58  has a length of about ¼ of a wavelength (λ) for the center frequency of interest. The gap  58  partially separates each element  52  into two lobes  60  which are connected at the outer extremities  68  of an element  52  and beyond the extent of the gap  58 . The lobes  60  of two adjacent elements  58  resemble to some extent a conventional Vivaldi notch antenna in that the edges  62  of the confronting, adjacent lobes  60  preferably assume the shape of a smooth departing curve. This shape of this curve can apparently be logarithmic, exponential, elliptic, or even be of some other smooth shape. The curves defining the edges  62  of adjacent lobes  60  diverge apart from the feed point  54 . The elements  52  are arranged about a center point  64  and their inner extremities  66  preferably lie on the circumference  69  of a circle centered on a center point  64 . The elements  52  extend in a generally outward direction from a central region generally defined by circumference  69 . The feed points  54  are also preferably located on the circumference of that circle and therefore each are located between (i) where the inner extremity  66  of one element  52  meets one of its edges  62  and (ii) where the inner extremity  66  of an adjacent element  52  meets its edge  62  which confronts the edge  62  of first mentioned element  52 . 
     The antenna  50  just described can conveniently be made using printed circuit board technology and therefore is preferably formed on an insulating substrate  88  (see FIG.  4 ). 
     Each element  52  is sized for the center frequency of interest. For example, if the antenna thus described were to be used for cellular communications services in the 1.8 Ghz band, then the length of the gap  58  in each element  52  is preferably about ¼ of a wavelength for the frequency of interest (1.8 Ghz in this example) and each element has a width of about 10 cm and a radial extent from its inner extremity  66  to its outer extremity  68  of about 11 cm. The antenna is remarkably wide banded and therefore these dimensions and the shape of the antenna can be varied as needed and may be adjusted according to the material selected as the insulating substrate and whether the antenna  50  is mounted adjacent a high impedance (Hi-Z) surface  70  (see FIGS.  3  and  4 ). The outer extremity  68  is shown as being rather flat in the figures, however, it may be rounded if desired. 
     Since the preferred embodiment has four elements  52  and since each pair of elements  52  forms a Vivaldi-like antenna we occasionally refer to this antenna as the Vivaldi Cloverleaf antenna herein, it being recognized that the Vivaldi Cloverleaf antenna can have fewer than four elements  52  or more than four elements  52  as a matter of design choice. 
     The Vivaldi Cloverleaf antenna  50  is preferably mounted adjacent a high impedance (Hi-Z) surface  70  as shown in FIGS. 3 and 4, for example. In prior art vehicular antennas the radiating structures are typically separated by at least one-quarter wavelength from nearby metallic surfaces. This constraint has severely limited where antenna could be placed on a vehicle and more importantly their configuration. In particular, prior art vehicular antennas tended to be non-aerodynamic in that they tended to protrude from the surface of the vehicle or they were confined to dielectric surfaces, such as windows, which often led to designs which were not particularly well suited to serving as omnidirectional antennas. 
     By following a simple set of design rules (see U.S. patent application Ser. No. 09/520,503 entitled “A Polarization Converting Radio Frequency Reflecting Surface” filed Mar. 8, 2000 mentioned above) one can engineer the band gap of the Hi-Z surface to prevent the propagation of bound surface waves within a particular frequency band. Within this band gap, the reactive electromagnetic surface impedance is high (&gt;377Ω), rather than near zero as it is for a smooth conductor. This allows antenna  50  to lie directly adjacent to the Hi-Z surface  70  without being shorted out as it would if placed adjacent a metal surface. The Hi-Z  70  may be backed by continuous metal such as the exterior metal skin of automobile, truck, airplane or other vehicle. The entire structure of the antenna  50  plus high impedance surface  70  is much thinner than the operating wavelength, making it low-profile, aerodynamic, and moreover easily integrated into current vehicle styling. Furthermore it is amenable to low-cost fabrication using standard printed circuit techniques. 
     Tests have been performed on a high impedance surface  70  comprising a three-layer printed circuit board in which the lowest layer  72  provides solid metal ground plane  73 , and the top two layers contain square metal patches  76 ,  82 . See FIGS. 5 and 6. The upper layer  80  is printed with 6.10 mm square patches  82  on a 6.35 mm lattice, which are connected to the ground plane by plated metal vias  84 . The second, buried layer  74  contains 4.06 mm square patches  76  which are electrically floating, and offset from the upper layer by one-half period. The two layers of patches were separated by 0.1 mm of polyimide insulator  78 . The patches in the lower layer are separated from the solid metal layer by a 5.1 mm substrate  79  preferably made of a standard fiberglass printed circuit board material commonly known as FR4. The pattern forms a lattice of coupled resonators, each of which may be thought of as a tiny LC circuit. In a geometry such as this, the proper unit for sheet capacitance is pF*square, and the proper unit for sheet inductance is nH/square. The overlap between the two layers of patches yields a sheet capacitance of about 1.2 pF*square, and the thickness of the structure provides a sheet inductance of about 6.4 nH/square. The resulting resonance frequency is:        f   =       1     2      π        LC         =     1.8                   GHz   .                                
     The width of the band gap can be shown to be:          f     Δ                 f       =           L   /   C             μ   o     /     ɛ   o           =     20        %   .                                
     To characterize the surface wave transmission properties of this high impedance, a pair of small coaxial probes were used. The last 1.5 cm of the outer conductor was removed from two pieces of semi-rigid coaxial cable, and the exposed center conductor acted as a surface wave antenna. The plot in FIG. 7 shows the surface wave transmission magnitude as a function of frequency. Between 1.6 and 2.0 GHz, a band gap is visible, indicated by the 30 dB drop in transmitted signal. Below the band gap, the surface is inductive, and supports TM surface waves, while above the band gap it is capacitive, and supports TE surface waves. Since the probes used in this experiment are much shorter than the wavelengths of interest, they tend to excite both TM and TE polarizations, so both bands can be seen in this measurement. For frequencies within the band gap, surface waves are not bound to the surface, and instead radiate efficiently into the surrounding space. An antenna  50  placed on such a surface will behave as though it were on an infinite ground plane, since any induced surface currents are forbidden from propagating by the periodic surface texture, and never reach the ground plane edges. An antenna  50  surrounded by a region of Hi-Z surface  70  can be placed arbitrarily on the metal exterior of a vehicle, with little variation in performance. Because of surface wave suppression, it will remain partially shielded from the effects of the surrounding electromagnetic environment, such as the shape of the ground plane. 
     The reflection phase of the surface was measured using a pair of horn antennas oriented perpendicular to the surface. Microwave energy is radiated from a transmitting horn, reflected by the surface, and detected with a receiving horn. The phase of the signal is recorded, and compared with a reference scan of a smooth metal surface, which is known to have a reflection phase of π. The reflection phase of the high impedance surface is plotted as a function of frequency in FIG.  8 . The surface is covered with a lattice of small resonators, which affect its electromagnetic impedance. Far below resonance, the textured surface reflects with a π phase shift, just as an ordinary metal surface does. Near resonance, the surface supports a finite tangential electric field across the capacitors, while the tangential magnetic field is zero, leading some to call this surface an artificial “magnetic conductor”. Far above resonance, the surface behaves as an ordinary metal surface, and the reflection phase approaches −π. Near the resonance frequency at 1.8 GHz, antenna  50  can be placed directly adjacent to the surface, separated by only a thin insulator  88  such as 0.8 mm thick FR4. The antenna  50  is preferably spaced a small distance (0.8 mm in this embodiment by the insulator  88 ) from the Hi-Z surface  70  so that the antenna  50  preferably does not interfere with the capacitance of the surface  70 . Because of the high surface impedance, the antenna is not shorted out, and instead it radiates efficiently. 
     Assuming that one pair of elements  52  are to be excited at any given time (when using the antenna  70  to transmit) or connected to a receiver at any given time (when using the antenna  70  to receive), then the four feed points  54 A,  54 B,  54 C and  54 D may be coupled to a radio frequency switch  90  (See FIG.  4 ), disposed adjacent the ground plane  73 , which switch  90  is coupled to the feed points  54 A,  54 B,  54 C and  54 D by short lengths  92  of a suitably shielded 50Ω cable or other means for conducting the radio frequency energy to and from the feed points through the Hi-Z surface  70  which is compatible with 50Ω signal transmission. By so connecting the antenna  50 , the RF switch  90  can be used to determine in which direction  56 A,  56 B,  56 C or  56 D the antenna  50  exhibits its highest gain by a control signal applied at control point  91 . The RF energy to and from the antenna is communicated via an RF port  93 . Alternatively, each feed point  54 A,  54 B,  54 C and  54 D can be coupled to demodulators and power meters for sensing the strength of the received signals before selecting the strongest signal by means of a RF switch  90 . 
     A test embodiment of the four adjacent elements  52 , which form the four flared notch antennas  53 , depicted by FIGS. 2 and 2 a  were disposed with their insulating substrate  88  on the test embodiment of the high impedance surface previously described with reference to FIGS. 5-8. The four antenna feed points  54 A,  54 B,  54 C and  54 D of the test embodiment were fed through the bottom of the Hi-Z surface  70  by four coaxial cables  92 , from which the inner and outer conductors are connected to the left and right sides of each feed point  54 . The four cables  92  were connected to a single feed by a 1×4 microwave switch  90  mounted below the ground plane  73 . In commercial embodiments a miniaturized version of this microwave switch could be attached to a recessed area in the center of the circuit board to further lower the antenna profile, if desired. The Hi-Z ground plane  70  for this test was 25.4 cm square while the breadth and width  67  of antenna  50  in this test embodiment measured 23.0 cm. Each flared notch gradually spread from 0.05 cm at the feed point  54  to 8.08 cm at the extremity of the antenna. In this test embodiment, the shape of the edges  62  of the lobes  60  was defined by an ellipse having major and minor radii of 11.43 cm and 4.04 cm, respectively. The isolating slots or gaps  58 , which are included to reduce coupling between adjacent elements  52 , had dimensions of 0.25 cm by 3.81 cm, and the circular central region  69  had a diameter of 2.54 cm. 
     To measure the radiation pattern, this test embodiment of antenna  50  with substrate  70  was mounted on a rotary stage, and the 1×4 RF switch  90  was used to select a single beam. The radiated power was monitored by a stationary horn as the test embodiment was rotated. Each of the four notch antennas  53  radiated a horizontally polarized beam directed at roughly 30 degrees above the horizon, as shown in the elevation pattern in FIG. 9. A 30-degree conical azimuth section of the radiation pattern was then taken by raising the receiving horn and scanning in the azimuth. The conical azimuth pattern of each flared notch antenna  53  covers a single quadrant of space as shown in FIG.  10 . The slight asymmetry of the pattern is due to the unbalanced coaxial feed. As such, some practicing the present invention want to elect to use a balanced feed instead However, we prefer an unbalance feed due to the simplicity gained by routing the signals to and from the antenna feed points  54  by means of coaxial cables. 
     The operating frequency and bandwidth of the antenna  50  are determined primarily by the properties of the Hi-Z surface  70  below it. The maximum gain of the antenna  50  occurred at a frequency of 1.8 GHz, near the resonance frequency of the Hi-Z surface. The gain decreased by 3 dB over a bandwidth of 10%, and by 6 dB over a bandwidth of 30%. In the elevation pattern, the angle of maximum gain varied from nearly vertical at 1.6 GHz to horizontal at 2.2 GHz. This is caused primarily by the fact that the Hi-Z surface  70  has a frequency dependent surface impedance. The azimuth pattern was more constant, and each of the four notch antennas  53  filled a single quadrant over a wide bandwidth. Specifically, the power at 45 degrees off the centerline  56  of a notch antenna  53  was between −3 and −6 dB of maximum over a range of 1.7 to 2.3 GHz. 
     FIG. 11 is a system diagram of a low profile, switched-beam diversity antenna system. The elements  52  of antenna  50  are shielded from the metal vehicle exterior  100  by a high impedance (Hi-Z) surface  70  of the type depicted by FIG. 1 a  or preferably a three layer Hi-Z surface as shown and described with reference to FIGS. 5-8. The total height of the antennas  50  and the Hi-Z surface  70  is much less than a wavelength (λ) for the frequency at which the antenna normally operates. The signal from each antenna feed point  54  is demodulated at a modulator/demodulator  20  using an appropriate input frequency or CDMA code  22  to demodulate the received signal into an Intermediate Frequency (IF) signal  24 . When the antenna  50  is used to transmit a RF signal, then the signal on line  29  is modulated to produce a transmitted signal. When the system of FIG. 11 is utilized as a receiver, then the power level of each IF signal  24  is then preferably determined by a power metering circuit  26 , and the strongest signal from the various sectors is selected by a decision circuit  28 . Decision circuit  28  includes a radio frequency switch  90  for passing the signal input and output to the appropriate feed point  54  of antenna  50  via an associated modem  20 . In this embodiment, a separate modulator/demodulator  20  is associated with each feed point  54 A,  54 B,  54 C and  54 D, although only two modulator/demodulators  20  are shown for ease of illustration. Correspondingly, the antenna  50  is shown in FIG. 11 as having two beams  1 , 2  associated therewith. Of course, the antenna shown in FIG. 2 would have four beam associated therewith, one for each feed point  54 . 
     Each pair of adjacent elements  52  of antenna  50  on the Hi-Z surface  70  form a notch antenna that has, as can be seen from FIG. 10, a radiation pattern that covers a particular angular section of space. Some pair of elements  52  may receive signals directly from a transmitter of interest, while others receive signals reflected from nearby objects, and still others receive interfering signals from other transmitters. Each signal from a feed point  54 A,  54 B,  54 C and  54 D is demodulated or decoded, and a fraction of each signal is split off by a signal splitter at numeral  23  to a separate power meter  25 . The output from the power meter  25  is used to trigger a decision circuit  27  that switches between the outputs  13  from the various demodulators. In the presence of multipath interference, the strongest signal is selected. In the presence of other interferers, such as other users on the same network, the signal  13  with the correct information is selected. In this case, the choice of desired signal is preferably determined by a header associated with each signal frame, which identifies an intended recipient. This task is preferably handled by circuitry in the modulator/demodulators. 
     The antenna  50  has a radiation pattern that is split into several angular segments. The entire structure can be very thin (less than 1 cm in thickness) and conformal to the shape of a vehicle, for example. The antenna  50  is preferably provided by a group of four flared notch antennas  53  arranged as shown in FIG.  4 . The antenna arrangement of FIG. 4 has been simulated using Hewlett-Packard HFSS software. The four rectangular slots or gaps  58  in the metal elements  52  are about one-quarter wavelength long and provide isolation between the neighboring antennas  53 . The importance of the slots has been shown in the simulations. The electric fields that are generated by exciting one flared notch antenna  53  are shown in FIG.  12 . The upper left quadrant is excited by a small voltage source at feed point  54 D and, as can be seen, the electric fields radiate outwardly along the flared notch section. They also radiate inwardly, along the edges of the circular central region  69 , but they encounter the rectangular slots  58  that effectively cancel out the currents. The result is a radiation pattern covering one quadrant of space, as shown in FIG.  13 . Exciting the other three feed points  54 A,  54 B,  54 C in a similar manner allows one to cover 360 degrees. More than four elements  52  could be provided to achieve finer beamwidth control. 
     The switched beam diversity and the High-Z surface technology discussed with reference to FIG. 11 does not necessarily depend on the use of a Vivaldi Cloverleaf antenna as the antenna employed in such as system. However, the use of the Vivaldi Cloverleaf antenna  50  has certain advantages: (1) it generates a horizontally polarized RF beam which (2) can be directionally controlled (3) without the need to physically re-orientate the antenna and (4) the antenna can be disposed adjacent to a metal surface such as that commonly found on the exteriors of vehicles. 
     If a vertically polarized beam is desired, then the wire antenna  50  shown in FIGS. 14 and 15 can be used in lieu of the Vivaldi Cloverleaf antenna  50 . Four wire antenna elements  52  are shown in FIG.  14 . Each element  52  is an elongated piece of wire having a feed point at one end thereof and having a length of more one than one half wavelength (0.5*λ) for the frequency of interest and less than one wavelength (λ) of the frequency of interest. Each wire antenna element  52  is preferably connected to an RF switch  90  and is disposed on a Hi-Z surface  70  with a thin intermediary layer  88  of polyimide, for example, disposed therebetween. 
     FIG. 16 is a graph of the elevation pattern of a beam radiated from a wire antenna element  52  disposed on the high impedance surface of FIGS. 5 and 6 while FIG. 17 is a graph of the radiation pattern taken through a 30 degree conical azimuth section of the beam transmitted from a wire antenna element  52  disposed on the high impedance surface of FIGS. 5 and 6. As can be seen this antenna is reasonably directional and therefore is a suitable choice for an antenna for use with the switched beam diversity system of FIG.  11 . 
     Other antenna geometries can provide finite directivity on a Hi-Z surface  70  and be suitable for use with the switched beam diversity system of FIG.  11 . 
     Having described this invention in connection with a preferred embodiment, modification will now certainly suggest itself to those skilled in the art. As such, the invention is not to be limited to the disclosed embodiments except as required by the appended claims.