Abstract:
A power factor correction circuit includes a filter capacitor interposed between AC input terminals, a first inductor interposed between the first end of the filter capacitor and an input of a first rectifying bridge circuit, and a second inductor interposed between the second end of the filter capacitor and another input of the first rectifying bridge circuit. The power factor correction circuit further includes a second rectifying bridge circuit including inputs connected to AC input terminals, and outputs connected to a smoothing capacitor. A control circuit controls switching devices in the first rectifying bridge circuit. The power factor correction circuit can facilitate preventing an overcurrent from flowing through diodes and parasitic diodes in the switching devices, using a simple configuration.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
       [0001]    This application claims priority under 35 U.S.C. §119 from JP PA 2009-133886, filed Jun. 3, 2009, the entirety of which is incorporated herein by reference. 
       FIELD OF THE INVENTION 
       [0002]    The present invention relates to a power factor correction circuit that controls the current flowing through AC input terminals by the ON-OFF operation of a switching device to be sinusoidal for obtaining a DC output. 
       BACKGROUND 
       [0003]      FIG. 17  is a circuit diagram of a conventional power factor correction circuit. Referring now to  FIG. 17 , the conventional power factor correction circuit conducts the full-wave-rectification of an AC voltage fed to AC input terminals  1   a  and  1   b  with rectifying bridge circuit  3  formed of diodes  3   a  through  3   d . The conventional power factor correction circuit boosts the output voltage fed from rectifying bridge circuit  3  with a boost chopper circuit formed of inductor  4 , MOSFET  6  working as a switching device, diode  8 , and smoothing capacitor  10 . Therefore, the DC voltage boosted by the boost chopper circuit described above is fed from DC output terminals  11   a  and  11   b.    
         [0004]    The ON-duty of MOSFET  6  in the boost chopper circuit is controlled so that the current flowing through AC input terminals  1   a  and  1   b  may be sinusoidal and so that the output from DC output terminals  11   a  and  11   b  may be constant. In  FIG. 17 , filter capacitor  2 , parasitic diode  6   a  of MOSFET  6  and bypass diode  15  for protecting MOSFET  6  and diode  8  from the rush current caused at the start of the circuit or at a power fail recovery are shown. 
         [0005]    In rectifying bridge circuit  3 , a current flows through two diodes without exception. In the state in which the voltage at AC input terminal  1   a  is higher than the voltage at AC input terminal  1   b , a current flows through diodes  3   a  and  3   d . In the state in which the voltage at AC input terminal  1   a  is lower than the voltage at AC input terminal  1   b , a current flows through diodes  3   b  and  3   c.    
         [0006]    Since the forward voltage drop caused when a current flows through any of diodes  3   a  through  3   d  is from 0.7 to 1 V, the total forward voltage drop caused when a current flows through diodes  3   a  and  3   d  or through diodes  3   b  and  3   c  is from 1.4 to 2 V. Therefore, as the capacity of the power supply to which the power factor correction circuit as described above is applied becomes larger, the losses caused in rectifying bridge circuit  3  become more pronounced, causing a lowering of conversion efficiency in the power factor correction circuit. 
         [0007]    Japanese Unexamined Patent Application Publication No. 2002-51563 (hereafter, “Patent Document 1”) and Japanese Unexamined Patent Application Publication No. 2004-72846 (hereafter, “Patent Document 2”) propose power factor correction circuits that provide the MOSFETs and diodes in the boost chopper circuit with a rectifying bridge function, for reducing the losses in rectifying bridge circuit  3 . 
         [0008]      FIG. 18  is a circuit diagram showing another conventional power factor correction circuit having a circuit configuration similar to the circuit configuration described in the Patent Document 1. The conventional power factor correction circuit shown in  FIG. 18  includes rectifying bridge circuit  12  including a series circuit of MOSFET  6  and diode  8  and a series circuit of MOSFET  7  and diode  9 . The series circuits are connected in parallel to each other. The first input of rectifying bridge circuit  12  is connected to AC input terminal  1   a  via inductor  4  and the second input thereof to AC input terminal  1   b . The first and second outputs of rectifying bridge circuit  12  are connected to DC output terminals  11   a  and  11   b , respectively. Filter capacitor  2  is interposed between AC input terminals  1   a  and  1   b . Smoothing capacitor  10  is interposed between DC output terminals  11   a  and  11   b . Parasitic diode  7   a  of MOSFET  7  is shown in  FIG. 18 . 
         [0009]    Now the operations of the circuit shown in  FIG. 18  will be described below with reference to  FIGS. 19A through 19D . 
         [0010]    As MOSFET  6  is turned ON in the state in which the voltage at AC input terminal  1   a  is higher than the voltage at AC input terminal  1   b , a current flows through the path described in  FIG. 19A  and an electric energy is stored in inductor  4 . A current flows through the path described in  FIG. 19A , as MOSFET  6  is turned OFF, and the electric energy stored in inductor  4  is released. On the current paths described in  FIGS. 19A and 19B , parasitic diode  7   a  of MOSFET  7  is present. 
         [0011]    As MOSFET  7  is turned ON in the state in which the voltage at AC input terminal  1   a  is lower than the voltage at AC input terminal  1   b , a current flows through the path described in  FIG. 19C  and an electric energy is stored in inductor  4 . A current flows through the path described in  FIG. 19D , as MOSFET  7  is turned OFF, and the electric energy stored in inductor  4  is released. On the current paths described in  FIGS. 19A and 19D , parasitic diode  6   a  of MOSFET  6  is present. 
         [0012]    In the circuit as shown in  FIG. 18 , the AC input voltage becomes higher than the voltage across capacitor  10  (the output voltage) sometimes. If the voltage at AC input terminal  1   a  is higher than the voltage at AC input terminal  1   b , when the AC input voltage is higher than the voltage across capacitor  10 , a rush current will flow through the path described in  FIG. 19B . If the voltage at AC input terminal  1   a  is lower than the voltage at AC input terminal  1   b , when the AC input voltage is higher than the voltage across capacitor  10 , a rush current will flow through the path described in  FIG. 19D . The occurrences of the rush currents imply that an overcurrent flows through diodes  8  and  9  and the insides (the parasitic diodes) of MOSFETs  6  and  7 . 
         [0013]    When MOSFET  6  or MOSFET  7  is turned ON in the circuit as shown in  FIG. 18 , diode  8  or diode  9  recovers reversely. Therefore, it is necessary to employ a high-speed diode that exhibits a short recovery time for diodes  8  and  9 . 
         [0014]    Since the high-speed diodes and the parasitic diodes in the MOSFETs (e.g. parasitic diodes  6   a  and  7   a  in  FIG. 18 ) exhibit a small current overload capacity, the high-speed diodes and the parasitic diodes may be broken down when the rush current is caused. 
         [0015]    Patent Document 2 proposes a power factor correction circuit as shown in  FIG. 20 , provided with a means for preventing the diodes and MOSFETs described above from being broken down. The circuit shown in  FIG. 20  includes diodes  3   a  and  3   b  and thyristors  14   a  and  14   b  added to the circuit shown in  FIG. 18 . Further, inductor  13  is connected in substitution for inductor  4  shown in  FIG. 18  for reducing the circuit size. 
         [0016]    In the power factor correction circuit shown in  FIG. 20 , diode  3   a  is interposed between AC input terminal  1   a  and DC output terminal  11   a , diode  3   b  between AC input terminal  1   b  and DC output terminal  11   a , thyristor  14   a  between AC input terminal  1   a  and DC output terminal  11   b , and thyristor  14   b  between AC input terminal  1   b  and DC output terminal  11   b.    
         [0017]    Inductor  13  includes inductors  13   a  and  13   b  coupled magnetically to each other. Inductor  13   a  is interposed between AC input terminal  1   a  and the first input of rectifying bridge circuit  12 . Inductor  13   b  is interposed between AC input terminal  1   b  and the second input of rectifying bridge circuit  12 . 
         [0018]    When a state in which an overcurrent flows through the path described in  FIG. 19B  is caused in the circuit shown in  FIG. 20 , thyristor  14   b  is turned ON before the overcurrent flows. By the turning ON of thyristor  14   b , the overcurrent is made to flow through diode  3   a  and thyristor  14   b , preventing diode  8  and MOSFET  7  from being broken down. 
         [0019]    When a state in which an overcurrent flows through the path described in  FIG. 19D  is caused in the circuit shown in  FIG. 20 , thyristor  14   a  is turned ON before the overcurrent flows. By the turning ON of thyristor  14   a , the overcurrent is made to flow through diode  3   b  and thyristor  14   a , preventing diode  9  and MOSFET  6  from being broken down. 
         [0020]    However, it is necessary to provide the circuit shown in  FIG. 20  separately with a power fail detector circuit (a circuit that detects the conditions under which an overcurrent is caused) that judges whether it is necessary to turn ON thyristor  14   a  or  14   b  or not, and with a driver circuit that drives thyristor  14   a  and  14   b . Therefore, the circuit configuration is complicated and the manufacturing costs of the circuit soar. If the power fail time is shorter than the time necessary for detecting the power fail, it will be impossible to accomplish the overcurrent bypass, since the power fail is not detected. 
         [0021]    In view of the foregoing, it would be desirable to obviate the problems described above. It would be also desirable to provide a power factor correction circuit that facilitates preventing an overcurrent from flowing through the high-speed diodes and the parasitic diodes in the MOSFETs with a simple circuit configuration, and that improves conversion efficiency and reliability. 
       SUMMARY OF THE INVENTION 
       [0022]    According to an aspect of the invention, there is provided a power factor correction circuit including a filter capacitor interposed between AC input terminals, and a first rectifying bridge circuit including a plurality of series circuits connected parallel to each other, each of the series circuits including a switching device and a rectifying device connected in series to each other. According to the aspect, the power correction circuit further includes a smoothing capacitor connected in parallel to the output of the first rectifying bridge circuit, a first inductor interposed between the first end of the filter capacitor and the first input of the first rectifying bridge circuit, and a second inductor interposed between the second end of the filter capacitor and the second input of the first rectifying bridge circuit. According to the aspect, the power correction circuit still further includes a second rectifying bridge circuit including diodes connected to form a bridge, the second rectifying bridge circuit including an input connected to any of the AC input terminals and an output connected to the smoothing capacitor, and a control circuit controlling the switching devices in the first rectifying bridge circuit. 
         [0023]    Advantageously, the first inductor and the second inductor are coupled magnetically to each other such that the directions of magnetic fluxes generated by the currents flowing through the first inductor and the second inductor are opposite to each other. 
         [0024]    Advantageously, the power factor correction circuit further includes another inductor or other inductors connected in series to either one or both of the first inductor and the second inductor. 
         [0025]    Advantageously, the first inductor and the second inductor are replaced by a leakage transformer including windings corresponding to the first inductor and the second inductor. 
         [0026]    Advantageously, the leakage transformer includes a core having a magnetic path leg positioned between the windings. 
         [0027]    Advantageously, the switching device in the first bridge circuit is a MOSFET. 
         [0028]    Advantageously, the control circuit includes an overcurrent detector and the control circuit suppresses an overcurrent, when the overcurrent is detected. 
         [0029]    According to another aspect of the invention, there is provided a power factor correction circuit including N groups of the power factor correction circuit described herein and connected in parallel to each other, wherein N=1, 2, 3, . . . . 
         [0030]    Advantageously, the ON-timings of the switching devices in the N groups of the power factor correction circuits are displaced by T/N, wherein T is the ON-period of the switching devices. 
         [0031]    Stated in other terms, embodiments of the invention relate to a circuit that can comprise a first bridge circuit including at least a switching device and a rectifying device connected in series, and a second bridge circuit including a plurality of diodes connected in series. The circuit can further comprise a first inductor connected between a first input of the first bridge circuit and a first input of the second bridge circuit, and a second inductor connected between a second input of the first bridge circuit and a second input of the second bridge circuit. 
         [0032]    The circuit can still further comprise a filter capacitor for connection to an AC input, wherein a first terminal of the filter capacitor is connected to the first input of the second bridge circuit, and a second terminal of the filter capacitor is connected to the second input of the second bridge circuit. The circuit can still further comprise a smoothing capacitor connected in parallel to an output of the first bridge circuit. Additionally, the circuit can yet further comprise an overcurrent detector, and a control circuit to suppress an overcurrent detected by the overcurrent detector. 
         [0033]    Embodiments of the invention further relate to a circuit that can comprise a first bridge circuit including at least a switching device and a rectifying device connected in series, and a second bridge circuit including a plurality of diodes connected in series. The circuit can further comprise an inductor including first and second windings wound around a core so as to be magnetically coupled. A first terminal of the first winding can be connected to a first input of the second bridge circuit, and a second terminal of the first winding can be connected to a first input of the first bridge circuit. A first terminal of the second winding can be connected to a second input of the second bridge circuit, and a second terminal of the second winding can be connected to a second input of the first bridge circuit. A direction of a magnetic flux generated by a current flowing from the first terminal of the first winding to the second terminal of the first winding can be opposite to a direction of a magnetic flux generated by a current flowing from the second terminal of the second winding to the first terminal of the second winding. 
         [0034]    The power factor correction circuit according to the invention exhibits the following effects. 
         [0035]    (1) The power loss caused in the diode in the second rectifying bridge circuit is reduced and the overcurrent caused at the start of operation of the circuit or at a power fail recovery is made not to flow through the diodes in the first rectifying bridge circuit (the high-speed diodes), nor through the parasitic diodes of the switching devices (the MOSFETs). 
         [0036]    (2) The changing rate of the current flowing through the inductor is reduced and the power losses caused in the switching devices and the rectifying devices are reduced. 
         [0037]    (3) The power losses caused in the inductors, switching devices and rectifying devices are dispersed. The downsizing of a heat sink for cooling the semiconductor devices and the downsizing of inductors and such component parts are facilitated. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0038]      FIG. 1  is a circuit diagram showing a power factor correction circuit according to a first embodiment of the invention. 
           [0039]      FIG. 2  is a circuit diagram describing the current path in the steady-state operations of the power factor correction circuit shown in  FIG. 1 . 
           [0040]      FIG. 3  is a circuit diagram describing the other current path in the steady-state operations of the power factor correction circuit shown in  FIG. 1 . 
           [0041]      FIG. 4  is a circuit diagram describing a current path in the non-steady-state operations of the power factor correction circuit shown in  FIG. 1 . 
           [0042]      FIG. 5  is a circuit diagram showing a power factor correction circuit according to a second embodiment of the invention. 
           [0043]      FIG. 6  is a drawing showing the detailed structure of the inductor shown in  FIG. 5 . 
           [0044]      FIG. 7  is a circuit diagram showing the current path in the steady-state operations in the power factor correction circuit according to the second embodiment. 
           [0045]      FIG. 8  is an equivalent circuit diagram of the power factor correction circuit according to the second embodiment that is operating as described in  FIG. 7 . 
           [0046]      FIG. 9  is a circuit diagram showing the other current path in the steady-state operations in the power factor correction circuit according to the second embodiment. 
           [0047]      FIG. 10  is a circuit diagram showing the current path in the non-steady-state operations in the power factor correction circuit according to the second embodiment. 
           [0048]      FIG. 11  is a schematic describing the structure of a leakage transformer. 
           [0049]      FIG. 12  is a schematic describing the structure of another leakage transformer. 
           [0050]      FIG. 13  is a circuit diagram showing a power factor correction circuit according to a third embodiment of the invention. 
           [0051]      FIG. 14  is a wave chart describing an example of the control mode for the MOSFET according to the third embodiment of the invention. 
           [0052]      FIG. 15  is a block circuit diagram showing an example of the control circuit that controls the ON and OFF of MOSFETs. 
           [0053]      FIG. 16  is a block circuit diagram showing another example of the control circuit that controls the ON and OFF of MOSFETs. 
           [0054]      FIG. 17  is a circuit diagram of a conventional power factor correction circuit. 
           [0055]      FIG. 18  is a circuit diagram of another conventional power factor correction circuit. 
           [0056]      FIG. 19A  is a first circuit diagram for describing the operations of the conventional power factor correction circuit shown in  FIG. 18 . 
           [0057]      FIG. 19B  is a second circuit diagram for describing the operations of the conventional power factor correction circuit shown in  FIG. 18 . 
           [0058]      FIG. 19C  is a third circuit diagram for describing the operations of the conventional power factor correction circuit shown in  FIG. 18 . 
           [0059]      FIG. 19D  is a fourth circuit diagram for describing the operations of the conventional power factor correction circuit shown in  FIG. 18 . 
           [0060]      FIG. 20  is a circuit diagram of still another conventional power factor correction circuit. 
       
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       [0061]    The invention will now be described in detail hereinafter with reference to the accompanying drawings. 
         [0062]      FIG. 1  is a circuit diagram showing a power factor correction circuit according to a first embodiment of the invention. In  FIG. 1 , the same reference numerals as used in  FIG. 18  are used to designate the same constituent elements and their duplicated description are omitted for the sake of simplicity. 
         [0063]    The power factor correction circuit according to the first embodiment is different from the conventional power factor correction circuit shown in  FIG. 18  in that rectifying bridge circuit  3  is interposed between AC input terminals  1   a  and  1   b  and rectifying bridge circuit  12 . The power factor correction circuit shown in  FIG. 1  is different from the conventional power factor correction circuit shown in  FIG. 18  also in that inductor  4   a  is interposed between AC input terminal  1   a  and the first input of rectifying bridge circuit  12 , and inductor  4   b  is interposed between AC input terminal  1   b  and the second input of rectifying bridge circuit  12 . In contrast, in the conventional power factor correction circuit shown in  FIG. 18 , only inductor  4  corresponding to inductor  4   a  is present. 
         [0064]    Rectifying bridge circuit  3  includes a bridge connection of diodes  3   a  through  3   d  as described earlier with reference to  FIG. 17 . Rectifying bridge circuit  12  includes a series circuit of MOSFET  6  and diode  8  and a series circuit of MOSFET  7  and diode  9 . The series circuits are connected in parallel to each other as described earlier with reference to  FIG. 18 . The inputs of rectifying bridge circuit  3  are connected to AC input terminals  1   a  and  1   b  and the output thereof to DC output terminals  11   a  and  11   b.    
         [0065]    Next, the operations of the power factor correction circuit according to the first embodiment will be described below with reference to  FIGS. 2 through 4 . 
         [0066]    First, the steady-state operations will be described. The steady-state operations include a first steady-state operation conducted when the voltage at AC input terminal  1   a  is higher than the voltage at AC input terminal  1   b , and a second steady-state operation conducted when the voltage at AC input terminal  1   a  is lower than the voltage at AC input terminal  1   b . Since the second steady-state operation can be estimated easily from the first steady-state operation, only the first steady-state operation will be described below. 
         [0067]    As MOSFET  6  is turned ON, a current flows through the path described in  FIG. 2 , and an electric energy is stored in inductor  4   a . Since the voltage applied across inductor  4   b  is the voltage difference between the forward voltage drop across diode  3   d  and the forward voltage drop across parasitic diode  7   a  of MOSFET  7 , the current that flows through inductor  4   b  (cf. the broken lines) is small. Therefore, most of the currents flow through the path described by the solid lines in  FIG. 2 . 
         [0068]    As MOSFET  6  is turned OFF, the electric energy stored in capacitor  4   a  is released through the path described in  FIG. 3 . Since the voltage applied across inductor  4   b  is the voltage difference between the forward voltage drop across diode  3   d  and the forward voltage drop across parasitic diode  7   a  of MOSFET  7 , the current that flows through inductor  4   b  (cf. the broken lines) is small. Therefore, most of the currents flow through the path described by the solid lines in  FIG. 3 . 
         [0069]    Next, the non-steady-state operations, conducted at the start of operation of the circuit or at a power fail recovery, will be described. The non-steady-state operations include a first non-steady-state operation conducted when the voltage at AC input terminal  1   a  is higher than the voltage at AC input terminal  1   b  and a second non-steady-state operation conducted when the voltage at AC input terminal  1   a  is lower than the voltage at AC input terminal  1   b . Since the second non-steady-state operation can be understood easily from the first non-steady-state operation, only the first non-steady-state operation will be described below. 
         [0070]    When the voltage across the AC input terminals is higher than the voltage across smoothing capacitor  10 , a current (a high rush current) that charges capacitor  10  flows through the path described in  FIG. 4 . Since the voltage applied across inductor  4   a  is the voltage difference between the forward voltage drop across diode  3   a  and the forward voltage drop across diode  8 , the current that flows through inductor  4   a  (cf. the single-dotted chain lines) is small. Since the voltage applied across inductor  4   b  is the voltage difference between the forward voltage drop across diode  3   d  and the forward voltage drop across parasitic diode  7   a  of MOSFET  7 , the current that flows through inductor  4   b  (cf. the broken lines) is small. 
         [0071]    As the above description clearly indicates, the power factor correction circuit according to the first embodiment makes the rush current (the charging current of capacitor  10 ) caused in the non-steady-state operations flow through the path described by the solid lines. Since any overcurrent does not flow through constituent elements  6  through  9  of rectifying bridge circuit  12 , constituent elements  6  through  9  are prevented from being broken by the overcurrent. The charging current of capacitor  10  flows through the diodes in rectifying bridge circuit  3 . Therefore, it is desirable to configure rectifying bridge circuit  3  with low-speed diodes for general rectification use that exhibit a high allowable peak current. 
         [0072]      FIG. 5  is a circuit diagram showing a power factor correction circuit according to a second embodiment of the invention. 
         [0073]    The power factor correction circuit according to the second embodiment is different from the power factor correction circuit according to the first embodiment in that inductors  4   a  and  4   b  are replaced by inductor  5  in  FIG. 5 . 
         [0074]      FIG. 6  is a drawing showing the detailed structure of inductor  5  shown in  FIG. 5 . 
         [0075]    As shown in  FIG. 6 , inductor  5  includes windings  5   a  and  5   b  wound around core  5   c  to be coupled magnetically to each other. Windings  5   a  and  5   b  are wound such that the direction of the magnetic flux ψ 1  generated by the current I 1  that flows from terminal A to terminal B is opposite to the direction of the magnetic flux ψ 2  generated by the current I 2  that flows from terminal C to terminal D. 
         [0076]    Next, the operations of the power factor correction circuit according to the second embodiment will be described below. 
         [0077]    First, steady-state operations will be described. The steady-state operations include a first steady-state operation conducted when the voltage at AC input terminal  1   a  is higher than the voltage at AC input terminal  1   b  and a second steady-state operation conducted when the voltage at AC input terminal  1   a  is lower than the voltage at AC input terminal  1   b . Since the second steady-state operation can be understood easily from the first steady-state operation, only the first steady-state operation will be described below. 
         [0078]    A current flows through the path described in  FIG. 7 , as MOSFET  6  is turned ON, and an electric energy is stored in the leakage inductance (not shown) of inductor  5 . Since the voltage across winding  5   b  is generated in the direction that blocks the current flowing through diode  3   d , the current that flows through diode  3   d  (cf. the broken lines) is small. Therefore, most of the currents flow through the path described by the solid lines. 
         [0079]    The operations described above will be described more in detail below. The equivalent circuit of the circuit described above in the operation mode described in  FIG. 7  is shown in  FIG. 8 . In  FIG. 8 , the self-inductance L 1  of winding  5   a , the self-inductance L 2  of winding  5   b , the mutual inductance M of the self inductance L 1  and the self inductance L 2 , the instantaneous value Vi of the AC input voltage, the current i 1  that flows through winding  5   a , the current i 2  that flows through winding  5   b , and the current i 3  that flows through diode  3   d  are shown. 
         [0080]    The relations described by the following differential equations hold from  FIG. 8 . 
         [0000]    
       
         
           
             
               
                 
                   
                     
                       
                         L 
                         1 
                       
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                     - 
                     
                       M 
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                             i 
                             2 
                           
                         
                         
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                           t 
                         
                       
                     
                   
                   = 
                   
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                   ( 
                   1 
                   ) 
                 
               
             
             
               
                 
                   
                     
                       
                         L 
                         2 
                       
                        
                       
                         
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                           t 
                         
                       
                     
                     - 
                     
                       M 
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                             i 
                             1 
                           
                         
                         
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                           t 
                         
                       
                     
                   
                   = 
                   0 
                 
               
               
                 
                   ( 
                   2 
                   ) 
                 
               
             
             
               
                 
                   
                     i 
                     3 
                   
                   = 
                   
                     
                       i 
                       1 
                     
                     - 
                     
                       i 
                       2 
                     
                   
                 
               
               
                 
                   ( 
                   3 
                   ) 
                 
               
             
           
         
       
     
         [0081]    The mutual inductance M in the equation (2) is expressed in the following way using the coupling coefficient k. 
         [0000]        M=k √{square root over ( L   1   L   2 )} 
         [0082]    By putting L 1  to be equal to L 2  and by rewriting the equation (2), the following equation (4) is obtained. 
         [0000]    
       
         
           
             
               
                 
                   
                     
                        
                       
                         i 
                         2 
                       
                     
                     
                        
                       t 
                     
                   
                   = 
                   
                     k 
                      
                     
                       
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                           i 
                           1 
                         
                       
                       
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                   ( 
                   4 
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         [0083]    From the equation (4), the relation expressed by the following equation (5) is obtained. 
         [0000]        i   2   =k·i   1   (5) 
         [0084]    From the equation (3), the relation expressed by the following equation (6) is obtained. 
         [0000]        i   3 =( 1 - k )· i   1   (6) 
         [0085]    As the equation (6) clearly indicates, the current i 3  that flows through diode  3   d  can be reduced by setting the coupling coefficient k at an appropriate value. 
         [0086]    As MOSFET  6  is turned OFF, a current flows through the path described in  FIG. 9  and the electric energy stored in the leakage inductance of inductor  5  is released. Since the voltage across winding  5   b  is generated in the direction that blocks the current flowing through diode  3   d , the current that flows through diode  3   d  (cf. the broken lines) is small. Therefore, most of the currents flow through the path described by the solid lines. 
         [0087]    The relation described by the foregoing equation (6) holds also in the operation mode described with reference to  FIG. 9 . Therefore, the current i 3  that flows through diode  3   d  can be reduced by setting the coupling coefficient k at an appropriate value. 
         [0088]    Next, the non-steady-state operations, conducted at the start of operation of the circuit or at a power fail recovery, will be described. The non-steady-state operations include a first non-steady-state operation conducted when the voltage at AC input terminal  1   a  is higher than the voltage at AC input terminal  1   b , and a second non-steady-state operation conducted when the voltage at AC input terminal  1   a  is lower than the voltage at AC input terminal  1   b . Since the second non-steady-state operation can be understood easily from the first non-steady-state operation, only the first non-steady-state operation will be described below. 
         [0089]    When the voltage across the AC input terminals is higher than the voltage across smoothing capacitor  10 , a current (a high rush current) that charges capacitor  10  flows through the path described in  FIG. 10 . Since the voltage applied across winding  5   a  of inductor  5  is the voltage difference between the forward voltage drop across diode  3   a  and the forward voltage drop across diode  8 , the current that flows through winding  5   a  (cf. the single-dotted chain lines) is small. Since the voltage applied across winding  5   b  of inductor  5  is the voltage difference between the forward voltage drop across diode  3   d  and the forward voltage drop across parasitic diode  7   a  of MOSFET  7 , the current that flows through winding  5   b  (cf. the broken lines) is small. 
         [0090]    As the above description clearly indicates, the power factor correction circuit according to the second embodiment makes most of the rush current (the charging current of capacitor  10 ) caused in the unsteady-state operations flow through the path described by the solid lines. Since any overcurrent does not flow through constituent elements  6  through  9  of rectifying bridge circuit  12 , constituent elements  6  through  9  are prevented from being broken by the overcurrent. 
         [0091]    Since the power factor correction circuit that employs inductor  5  facilitates reducing the number of the inductors as compared with the power factor correction circuit shown in  FIG. 1 , the power factor correction circuit according to the second embodiment is more advantageous to reduce the size and the manufacturing costs of the power factor correction circuit. 
         [0092]    The charging current of capacitor  10  flows through the diodes in rectifying bridge circuit  3 . Therefore, it is desirable to configure rectifying bridge circuit  3  with low-speed diodes for general rectification use that exhibit a high allowable peak current. The current that flows through rectifying bridge circuit  3  in the steady state operations is small. The charging current that charges capacitor  10  (the rush current) at the start of operation of the circuit or at a power fail recovery flows through rectifying bridge circuit  3  transiently for a short time. If one considers these facts, the rated current of rectifying bridge circuit  3  may be set to be small with no problem. 
         [0093]    When the leakage inductance of inductor  5  is small, the changing rate of the current that flows through inductor  5  is large. In this case, the turnoff currents of MOSFETs  6  and  7  become high, increasing the turnoff losses and the conduction losses. When there is a concern that the turnoff losses and the conduction losses will be too large, another inductor or other inductors are connected to either one or both of windings  5   a  and  5   b  in series to lower the changing rate of the current that flows through inductor  5 . 
         [0094]    For obtaining much larger leakage inductance, inductor  5  may be provided with the leakage transformer structures as shown in  FIGS. 11 and 12 . 
         [0095]    It is possible for the leakage transformer having the structure shown in  FIG. 11  to increase the leakage inductance by adjusting the spacing d between windings  5   a  and  5   b . It is possible for the leakage transformer having the structure shown in  FIG. 12  to increase the leakage inductance by providing core  5   c  with a magnetic path leg positioned between windings  5   a  and  5   b  and by adjusting the gap length g formed by the magnetic path leg. 
         [0096]    If inductor  5  is provided with any of the structures described above to obtain large leakage inductance, it will be possible to lower the changing rate of the current flowing through the inductor  5  without connecting another inductor or other inductors to either one or both of windings  5   a  and  5   b  in series. 
         [0097]      FIG. 13  is a circuit diagram showing a power factor correction circuit according to a third embodiment of the invention. The power factor correction circuit according to the third embodiment includes two groups of the circuit shown in  FIG. 5  and connected in parallel to each other. The constituent elements in the added group of the circuit are marked with a prime “‘ ” in  FIG. 13 . 
         [0098]    The power factor correction circuit according to the third embodiment facilitates reducing the electric power transmitted through one group of the circuit as compared with the circuit shown in  FIG. 5 . Therefore, the losses caused by inductors  5 ,  5 ′, MOSFETs  6 ,  6 ′,  7 ,  7 ′, and diodes  8 ,  8 ′,  9 ,  9 ′ are dispersed, and, as a result, the downsizing of the heat sink for cooling the semiconductor devices (MOSFETs  6 ,  6 ′,  7 ,  7 ′, etc.) and the downsizing of inductors  5  and  5 ′ are facilitated. 
         [0099]    The circuit, a plurality of which is connected in parallel, is not always limited to the circuit shown in  FIG. 5 . The circuit, a plurality of which is to be connected in parallel, may be the circuit shown in  FIG. 1  with no problem. The number of groups connected in parallel is not always limited to 2. Three or more groups of the circuit may be connected in parallel with no problem. In the power factor correction circuit shown in  FIG. 13 , rectifying bridge circuit  3  and smoothing capacitor  10  are used commonly for the circuit groups. Alternatively, each of the circuit groups may be provided with rectifying bridge circuit  3  and/or smoothing capacitor  10  individually. 
         [0100]    The gate signals described in  FIG. 14  are fed to MOSFETs  6  and  6 ′, respectively. The gate signals of MOSFETs  6  and  6 ′ are generated for the period T and the timings thereof are displaced by 0.5 T from each other. 
         [0101]    If MOSFETs  6  and  6 ′ are controlled by the gate signals as described above, MOSFETs  6  and  6 ′ are ON for the periods Ton and Ton′, respectively. Therefore, the ripple of the synthesized current Iac of the currents IL 1  and IL 2  flowing through inductors  5  and  5 ′ is minimized and the size of filter capacitor  2  and the size of the line filters (not-shown) connected to AC input terminals  1   a  and  1   b  are reduced. MOSFETs  7  and  7 ′ are controlled in the same manner as MOSFETs  6  and  6 ′. 
         [0102]    If the ON-timings of the switching devices (the MOSFETs) in the adjacent groups are displaced from each other by T/N, in which N (N=2, 3, 4, . . . ) is the number of the power factor correction circuit groups connected in parallel, the effects described above (the effect of reducing the electric power transmitted through each circuit group and the effect of reducing the ripple of the synthesized current) are enhanced more as N is larger. 
         [0103]    Although not illustrated in  FIGS. 1 and 5 , MOSFETs  6  and  7  in rectifying bridge circuit  12  in the power factor correction circuits shown in these drawings are controlled by the control circuit as shown in  FIG. 15 . Control circuit  100  shown in  FIG. 15  is applied to the circuit shown in  FIG. 1 . Control circuit  100  shown in  FIG. 15  is applicable also to the circuit shown in  FIG. 5 . 
         [0104]    Control circuit  100  includes reference voltage source  100   a , voltage error amplifier  100   b , current error amplifier  100   c , multiplier  100   d , absolute value circuits  100   e  and  100   e ′, PWM comparator  100   f , PWM carrier signal generator circuit  100   g , and current detector  200 . 
         [0105]    Voltage error amplifier  100   b  amplifies the difference between a feedback signal corresponding to the output voltage (the voltage across smoothing capacitor  10 ) and a reference voltage fed from reference voltage source  100   a . Voltage error amplifier  100   b  feeds a first error signal corresponding to the amplified difference. The feedback signal includes the detected but not modified output voltage, the divided value of the detected output voltage, and the detected output voltage, the level of which has been shifted. 
         [0106]    Multiplier  100   d  multiplies the first error signal and the absolute value of the signal corresponding to the input voltage (the voltage across filter capacitor  2 ) and feeds the result of the multiplication to current error amplifier  100   c  as a reference current value. The signal corresponding to the input voltage includes the detected but not modified input voltage, the divided value of the input voltage, and input voltage, the level of which has been shifted. 
         [0107]    Current error amplifier  100   c  amplifies the difference between the multiplication result described above and the absolute value of a signal corresponding to the current (the output current) detected by current detector  200  and feeds a second error signal corresponding to the amplified difference. 
         [0108]    PWM comparator  100   f  compares the second error signal and a carrier signal such as a triangular wave signal and a saw-tooth wave signal fed from PWM carrier signal generator circuit  100   g  and feeds a PWM signal having a duty ratio corresponding to the magnitude of the second error signal. 
         [0109]    Overcurrent detector circuit  100   j  feeds an overcurrent detection signal at a low level (hereinafter referred to as an “L-level”), when the absolute value of the output current detected by current detector  200  exceeds a predetermined value to the higher side. 
         [0110]    AND circuit  100   k  calculates the logic AND of the PWM signal and the output signal from overcurrent detector circuit  100   j  and feeds the result of the AND calculation to the gates of MOSFETs  6  and  7  via gate drivers  100   h  and  100   i . Therefore, MOSFETs  6  and  7  are turned ON and OFF simultaneously by the same gate signal. As overcurrent detector circuit  100   j  detects an overcurrent, both MOSFET  6  and MOSFET  7  are turned OFF. 
         [0111]    Control circuit  100  facilitates controlling the ON and OFF of MOSFETs  6  and  7  based on the input voltage (the voltage across filter capacitor  2 ), the output voltage (the voltage across smoothing capacitor  10 ), and the output current such that the current flowing through AC input terminals  1   a  and  1   b  is shaped with a sinusoidal wave and such that the output from DC output terminals  11   a  and  11   b  is constant (equal to a reference voltage). 
         [0112]    Control circuit  101  shown in  FIG. 16  may be employed in substitution for control circuit  100 . Control circuit  101  is different from control circuit  100  in that control circuit  101  includes comparator  101   a , NOT circuit  101   b , OR circuits  101   c  and  101   d , and AND circuits  101   e  and  101   f.    
         [0113]    Comparator  101   a  determines the polarity of the input voltage (the voltage across filter capacitor  2 ) and feeds an L-level signal, when the voltage at the input terminal  1   a  is positive, or a high level (hereinafter referred to as an “H-level”) signal, when the voltage at the input terminal  1   b  is positive. 
         [0114]    OR circuit  101   c  calculates the logic OR of the PWM signal fed from PWM comparator  100   f  and the output signal fed from comparator  101   a . AND circuit  101   e  calculates the logic AND of the output signal from OR circuit  101   c  and the output signal from overcurrent detector circuit  100   j  and feeds the result of the AND calculation to the gate of MOSFET  6  via gate driver  100   h.    
         [0115]    OR circuit  101   d  calculates the logic OR of the PWM signal and the output signal from NOT circuit  101   b  connected to the output of comparator  101   a . AND circuit  101   f  calculates the logic AND of the output signal from OR circuit  101   d  and the output signal from overcurrent detector circuit  100   j  and feeds the result of the AND calculation to the gate of MOSFET  7  via gate driver  100   i.    
         [0116]    Control circuit  101  controls MOSFETs  6  and  7  in the different manner as described below depending on the polarity of the input voltage (the voltage across filter capacitor  2 ). 
         [0117]    When the voltage of input terminal  1   a  is positive: 
         [0118]    Comparator  101   a  feeds an L-level signal. Therefore, OR circuit  101   c  feeds the PWM signal described above and OR circuit  101   d  sets the output thereof at the H-level. As a result, if overcurrent detector circuit  100   j  has not detected any overcurrent, the ON and OFF of MOSFET  6  will be controlled based on the PWM signal described above and MOSFET  7  will maintain the ON-state thereof. 
         [0119]    When the voltage of input terminal  1   b  is positive: 
         [0120]    Comparator  101   a  feeds an H-level signal. Therefore, OR circuit  101   d  feeds the PWM signal described above and OR circuit  101   c  sets the output thereof at the H-level. As a result, if overcurrent detector circuit  101   j  has not detected any overcurrent, MOSFET  6  will maintain the ON-state thereof and the ON and OFF of MOSFET  7  will be controlled based on the PWM signal described above. 
         [0121]    If overcurrent detector circuit  100   j  has detected an overcurrent, MOSFETs  6  and  7  are both turned OFF in the same manner as in the foregoing control circuit  100 . 
         [0122]    The control circuit that generates the gate signals as described in  FIG. 14  may be provided with the structure shown in  FIG. 15  or  16 . Note that it is necessary for the control circuit described above to be provided additionally with a delay element to displace the ON-timings of MOSFETs  6  and  6 ′ (MOSFETs  7  and  7 ′) by T/N. 
         [0123]    Although the invention has been described in connection with the embodiments thereof, changes and modifications are obvious to the persons skilled in the art without departing from the true spirit of the invention. 
         [0124]    For example, the power factor correction circuits according to the invention have been described in connection with a single-phase AC input voltage, the power factor correction circuit according to the invention may be configured for a three-phase AC input voltage. In the modification, rectifying bridge circuits  3  and  12  and inductors  4   a ,  4   b , and  5  are replaced by the respective alternatives having the respective structures suitable for a three-phase AC input voltage. 
         [0125]    It will be apparent to one skilled in the art that the manner of making and using the claimed invention has been adequately disclosed in the above-written description of the exemplary embodiments taken together with the drawings. Furthermore, the foregoing description of the embodiments according to the invention is provided for illustration only, and not for limiting the invention as defined by the appended claims and their equivalents. 
         [0126]    It will be understood that the above description of the exemplary embodiments of the invention are susceptible to various modifications, changes and adaptations, and the same are intended to be comprehended within the meaning and range of equivalents of the appended claims.