Abstract:
A tunable bandpass filter integrated circuit includes a filter core including at least two spaced conductor layers, a plurality of peripherally spaced backside vias extending between the conductor layers defining a resonator cavity, at least one internal backside via, and a tunable impedance connected in series with the internal backside via between the conductor layers for adjusting the resonance of the cavity.

Description:
FIELD OF THE INVENTION 
     This invention relates to a tunable bandpass filter integrated circuit, and more particularly to one which may be embodied in MMIC using a varactor or similar tuning device and which is operational at and above millimeter wavelength. 
     BACKGROUND OF THE INVENTION 
     Microstrip and stripline techniques are very popular at microwave frequencies due to their small physical size, versatility, price and ease of construction. However at millimeter wave frequencies and above, problems occur that limit the usability of such structures. The wavelengths become so small that parasitic effects brought by practical sized interconnections, such as input/output coupling and intra-layer transition vias are performance limiting, and unintended modes occur in the design. In order to reduce these effects one has to go to smaller and smaller layer thicknesses, which causes a reduction in the widths of the microstrips/striplines and exacerbating the natural metal losses of the resonators that increase with frequency. For these reasons microstrip/stripline based resonator structures can become too lossy for certain applications. 
     Mechanical filters offer distinct advantages at the above mentioned frequencies. The use of large metal surface resonators circumvents the electrical loss issue and coupling into and between resonators can be done through openings in the relatively large cavity resonators. However, these structures are large, hard to integrate with other components and extremely expensive 
     Laminate based technology has been used in the past, see U.S. Pat. Nos. 6,535,083, 6,137,383 incorporated by reference herein. See also U.S. Pat. Nos. 5,821,836, 6,362,706, 6,535,083, and 5,382,931 which disclose constructing combline bandpass filter structures incorporated by reference herein. Walls formed by plated through holes or vias define dielectric filled waveguide structures. Furthermore, plated through holes that do not go all the way through and are situated inside the structures are used as the combline resonator elements. These prior art devices are fixed frequency filters and are not tunable unless separate non-integrated elements are added which negatively affect cost and performance. Integration of separate tuning elements is possible but limitations in miniaturization and parasitics will prevent high frequency operation. 
     An electrically tunable filter provides great flexibility in system architectures, by being able to replace multiple fixed frequency filters and switch matrices but this can result in interconnection parasitics between filter resonators and tunable elements at these frequencies. 
     SUMMARY OF THE INVENTION 
     It is therefore an object of this invention to provide an improved tunable bandpass filter integrated circuit. 
     It is a further object of this invention to provide such an improved tunable bandpass filter integrated circuit which is electrically tunable. 
     It is a further object of this invention to provide such an improved tunable bandpass filter integrated circuit which is implementable in fully integrated technology such a MMIC. 
     It is a further object of this invention to provide such an improved tunable bandpass filter integrated circuit which is applicable at and above mmW frequencies. 
     It is a further object of this invention to provide such an improved tunable bandpass filter integrated circuit which is relatively simple and inexpensive to implement. 
     The invention results from the realization that a tunable bandpass filter integrated circuit can be achieved using a filter core including at least two spaced conductor layers with a plurality of peripherally spaced backside vias extending between the conductor layers to define a resonator cavity, at least one internal via and a tunable impedance connected in series with the internal backside via between the conductor layers for adjusting the resonance of the cavity. 
     This invention features a tunable bandpass filter integrated circuit including a filter core which includes at least two spaced conductor layers, a plurality of peripherally spaced backside vias extending between the conductor layers defining a resonator cavity, at least one internal backside via, and a tunable impedance connected in series with the internal backside via between the conductor layers for adjusting the resonance of the cavity. 
     In a preferred embodiment the filter core may include a semiconductor material. The bandpass filter integrated circuit may be a microwave monolithic integrated circuit (MMIC). The bandpass filter integrated circuit may operate at and above millimeter wavelengths (mmW). The semiconductor material may include a low conductivity silicon. The semiconductor material may include gallium arsenide. The conductor planes may be on the top and bottom of the core. The tunable impedance may include a varactor. The tunable impedance may include a MEMS device. The tunable impedance may include a ferroelectric dielectric. At least one of the peripherally spaced backside vias may be omitted in at least two locations to form input and output ports. There may be two separated internal backside vias each establishing a resonator and each connected in series with a respective tunable impedance between the at least two conductor layers and a secondary metallization member extends between each of the tunable filters and one of the input and output ports. There may be a plurality of separated internal backside vias each establishing a resonator and each connected in series with a respective tunable impedance between the at least two conductor layers. Each pair of resonators may constitute a simple filter. The backside vias separating adjacent resonators may be omitted establishing an evanescent mode waveguide. There may be between adjacent resonators an opening in a conductor layer with a bridging secondary metallization element for coupling between those adjacent resonators. The extremities of the bridging secondary metallization element may be short circuited at the frequency of operation. There may be an inter-resonator tunable capacitance between resonators for controlling inter-resonator coupling. There may be coupling between non-adjacent resonators to establish asymmetrical bandpass response with one or more finite frequency nulls. The resonators may be in a line. The resonators may be in a folded path. The tunable impedance may include back-to-back connected varactors for mitigating large signal distortions. The back-to-back connected varactors may include a resistance for controlling the slope of amplitude equalization. The integrated circuit may include bump pads for flip chip mounting. 
     The subject invention, however, in other embodiments, need not achieve all these objectives and the claims hereof should not be limited to structures or methods capable of achieving these objectives. 
    
    
     
       BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS 
       Other objects, features and advantages will occur to those skilled in the art from the following description of a preferred embodiment and the accompanying drawings, in which: 
         FIG. 1  is a three dimensional diagrammatic view of a cavity resonator with backside vias; 
         FIG. 2  is a view similar to  FIG. 1  with filter core removed for clarity; 
         FIG. 3  is a schematic block diagram of a bandpass filter comprised of coupled resonators; 
         FIG. 4  is an LC equivalent circuit model of a cavity resonator; 
         FIG. 5  is a view similar to  FIG. 1 , of a resonator with an internal backside via according to this invention; 
         FIG. 6  is a view similar to  FIG. 1  of a resonator with an internal via and capacitors with two ports; 
         FIG. 7  is an equivalent circuit of the resonator of  FIG. 6  in the nature of a two port comb resonator; 
         FIG. 8  is a view similar to  FIG. 1  of a two cavity combline filter operating in a combline mode; 
         FIG. 9  is a view similar to  FIG. 8  with the top conductor removed for clarity; 
         FIGS. 10A-D  are schematic diagrams of four different examples of transmission lines using secondary metallization at the “top” conductor; 
         FIG. 11  is a view similar to  FIG. 1  of a multiple combline resonator in an evanescent mode coupled filter; 
         FIG. 12  shows a portion of multiple combline resonator similar to that of  FIG. 11  with coupling through secondary metallization; 
         FIG. 13  is a view similar to  FIG. 12  where the coupling includes a variable capacitance device; 
         FIG. 14  is a view similar to  FIG. 1  of a multiple combline resonator with non-adjacent coupling with the top conductor mode transparent for clarity; 
         FIG. 15  is a view similar to  FIG. 14  but with the structure folded; 
         FIG. 16  is a view of  FIG. 15  with the top conductor removed for clarity; 
         FIG. 17  is a view similar to  FIG. 16  with non-adjacent coupling between first and last resonators yielding asymmetrical electrical bandpass response; 
         FIG. 18  is an equivalent circuit of a five resonator coupled filter; 
         FIG. 19  is a representation of the transfer function of the filter of  FIG. 18 ; 
         FIG. 20  is a view similar to  FIG. 18  with additional equalization resistances; and 
         FIG. 21  illustrates the use with resonator filters of bonding bumps for flip-chip bonding. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Aside from the preferred embodiment or embodiments disclosed below, this invention is capable of other embodiments and of being practiced or being carried out in various ways. Thus, it is to be understood that the invention is not limited in its application to the details of construction and the arrangements of components set forth in the following description or illustrated in the drawings. If only one embodiment is described herein, the claims hereof are not to be limited to that embodiment. Moreover, the claims hereof are not to be read restrictively unless there is clear and convincing evidence manifesting a certain exclusion, restriction, or disclaimer. 
     This invention in one embodiment provides a voltage control filter that incorporates structures to define by backside vias in microwave monolithic integrated circuits (MMIC). Bandpass filter responses at frequencies in the millimeter wavelength (mmW) range and higher can be obtained. Backside vias are used in MMIC&#39;s to create walls whereby electromagnetic energy can be confined to create electrical resonant cavities, see U.S. Provisional Patent Ser. No. 61/572,320 filed Jul. 14, 2011 incorporated by reference herein. Such a resonant cavity  10  is shown in  FIG. 1 , as including core  12  and two spaced conductor layers  14  and  16  electrically interconnected by backside vias  18 . Core  10  is shown again in  FIG. 2 , with the core  12  removed for clarification. 
     At very high frequencies excessive filter loss is avoided and out of band suppression is increased by using a cascade of very high Q resonant cavities as shown in  FIG. 3  where resonant cavities Q,  20 ,  22 ,  24  and  26  are coupled to each other by constant circuits K  28 ,  30  and  32  and the cavities at each edge are coupled externally with circuits Kin,  34  and  36 . The size of the mmW frequency resonant cavity within an MMIC is on the order of several millimeters. A bandpass filter consisting of multiple resonant cavities fabricated with MMIC technology can be large and expensive. Additional elements can be inserted such that the resonant frequency can be altered, as described in see U.S. Provisional Patent Ser. No. 61/572,320 filed Jul. 14, 2011 incorporated by reference herein. These additional elements can be used reduce the size of the resonators such that the cavity resonator based bandpass filters become feasible within an MMIC. The “K” circuits represent the coupling between resonant cavities. The equivalent electrical circuit of the resonant cavity  20 - 26  is shown in  FIG. 4  where capacitance  40 , C res  and inductance  42 , L res  represent the inductance capacitance of the cavity resonator. 
     In contrast resonant cavity frequency can be increased by the addition of one or more backside vias such as internal via  44  in generally the middle region of the resonant cavity as shown in  FIG. 5 . In subsequent figures, like parts have been given like numbers and similar parts have been given like numbers accompanied by a lower case letter. Core  12   a  may be a semi-conductor material with dielectric properties such as gallium arsenide or silicon and may include other materials such as thin layers of dielectric as in  FIG. 10B . Spaced conductor plates  14   a  and  16   a  are shown on the top and bottom surfaces of core  12   a  but this is not a necessary limitation. In other embodiments of the invention, in order to reduce the resonant frequency of the cavity  10   a , the internal backside via or vias can have electrically capacitive tuning elements such as varactors, or MEMS, or ferroelectric dielectric materials  50 ,  FIG. 6 , attached to them at the point where they meet the top metallization or conductor  14   b  such as reverse biased collector-base diodes, micro electro mechanical plates and barium strontium titanate ferroelectric dielectric. In  FIG. 6  at least one backside via has been removed at each end to create ports  52  and  54  where diodes  53  and  55  represent the impedance of the port. The equivalent circuit of such a two part comb resonator is shown in  FIG. 7  where the inductive resonances  56  and  58  at ports  52  and  54 , respectively, are much higher than the general inductance of the resonator L RES    60 , the capacitance of the resonator C RES  is represented at  62 , the capacitance C via  at  63  and inductance L via  is shown at  64 , respectively. Coupling into the bandpass filter can be done using any of the techniques shown in see U.S. Provisional Patent Ser. No. 61/572,320 filed Jul. 14, 2011 incorporated by reference herein. 
     A combline filter  70 ,  FIG. 8 , operational in combline mode is shown with peripheral back side vias removed at areas  72  and  74  to create ports  52   c  and  54   c , respectively and removed at area  76  to create an intermediate port  78 . Combline filter  70  is shown with somewhat more clarity in  FIG. 9 , with the top conductor  14   c  removed. Stronger input coupling is provided in combline filter  70  by using secondary metallization  80  and  82  which runs underneath the top metallization layer or conductor  14   c  with the two forming electromagnetic transmission lines and connecting between each of the inner/central backside vias  44   c  and the variable capacitors  50   c  of the respective cavity resonators  10   e . In  FIG. 9  resonators  10   c  form a simple filter  70 . The filter center frequency is adjusted by a backside via  44   c  located internally and preferably in the middle of each resonator, the variable capacitors and the cavity dimensions. The coupling between the two resonators in  FIG. 9  is adjusted by the cavity dimensions, the distance between the two resonators and the opening of the backside via defined inner wall between resonators. 
     There are a number of different ways that transmission lines may use the secondary metallization with the top metal, for example. Four such ways are shown in  FIGS. 10A , B, C, D. In  FIG. 10A  there is a dielectric  86  on top of the top metal  14   d  and the secondary metallization  80   d  is on top of the dielectric. In  FIG. 10B  the dielectric  86  is under the top metal  14   d  and forms a part of the cord  12   d . The secondary metallization is under the dielectric. Dielectric  86  may include SiO 2 , SiN or similar materials. In  FIG. 10C  the dielectric  86  is on top of the top metal  14   d  and the secondary metallization  80   d  is on top of the dielectric similar to  FIG. 10A  but there is a gap  88  in top metal  14   d  proximate secondary metallization  80   d . In  FIG. 10D  there is also a gap in the top metal  14   d  but the secondary metallization  80   d  is disposed in gap  88  and spaced from the edges of dielectric  14   d  defined by gap  88 . 
     In accordance with this invention a filter can be formed as with the single resonator  10   b  as shown in  FIG. 6  or in another preferred embodiment it can be formed by multiple resonators coupled to each other as in  FIG. 11  where there are four resonators  10   e , which are essentially multiple combline resonators forming an evanescent mode coupled filter  70   e . The coupling between resonators  10   e  can be adjusted by the opening left by the omitted backside vias in the areas separating the adjacent resonators  10   e . When all of the backside vias between cavities are removed as in  FIG. 11  an evanescent mode waveguide structure is obtained.  FIG. 11  is, in fact, a realistic implementation of the block diagram of  FIG. 3 . Secondary metallization  80   e  and  82   e  in  FIG. 11  that is just beneath the top metallization is connected to the first and last resonators  10   e  to couple in and out of filter  70   e . Inter-resonator coupling can also be adjusted using secondary metallization situated beneath or above the top surface as indicated previously. 
       FIG. 12  illustrates inter-resonator coupling using secondary metallization  89  situated above the top metal  14   f  of the cavity  10   f . Here slots  90 ,  92  are used to couple energy into transmission line  89  and then back into the next resonator. The secondary metal  89  in this example could just as well have been below top metal  14   f . The extremities of line  89  can be terminated in a number of ways and for greater coupling they can be effectively short circuited at the frequency of operation such as by shorting blocks  93 ,  95  which extend through insulating layer  97  to conductor layer  14   f . That short circuit can be a quarter wavelength extension, a large valued capacitor or a physical short as shown in  FIG. 12 . Since backside via placement tolerances can be relatively high, this technique is beneficial when low levels of coupling between resonators need to be well controlled. Although there may be various layers associated with conductor layers  14  and  16  and core  12 , they have been omitted for clarity generally in the figures except where needed for understanding. 
     The level of inter-resonator coupling can be electrically adjusted by inserting tunable capacitors  100 ,  FIG. 13 , in the top metallization  14   g ,  FIG. 13 , or inserted in the path of the secondary metallization line  89  of  FIG. 12 , for example. That secondary line  89  of  FIG. 12  can be modified in a number of ways in order to vary the coupling amplitude and phase versus frequency characteristics. 
     When resonators are strictly coupled only to adjacent resonators input to output electrical response will have a bandpass characteristic, but it will not have a transmission zero at finite frequency. The rejection of the filter will increase as the frequency tends to infinity and to zero. However when signals are coupled from one resonator  10   h  to another,  FIG. 14  at specific magnitude and phase into non-adjacent resonators asymmetrical bandpass responses are obtained where finite frequency nulls are present. The non-adjacent coupling can be accomplished by additional secondary metallization members  102  and  104 , for example, in  FIG. 14 . Thus the energy can be picked up from one resonator and routed through the walls formed by the backside vias  18  and then delivered to any of the other resonator in the filter. In  FIG. 14  the signal is picked up from the first resonator  10   h , delivered through secondary metallization  102  and  104  and into the last resonator  10   h .  FIGS. 15 and 16  illustrate another implementation where the cavities  10   i   1 - 10   i   4  are not in a straight line but are folded as indicated by arrow direction line  110  so that filter  70   i  becomes more compact and the input/output sections  52   i ,  54   i  are next to each other. In this structure an asymmetrical bandpass response can easily be created since resonant structures  10   i   1  and  10   i   4  are side by side and the required low level coupling is easily delivered through secondary metallization such as line  112 . In  FIG. 17  a backside via has been removed to create the coupling between non-adjacent resonant cavities in the filter by the removal of one or more backside vias in the area indicated at  114 . 
     The distortion created when large signal levels are applied is improved by incorporating back to back (cathode to cathode or anode to anode) pairs of varactors for each single varactor in the filter substantially eliminating the non-symmetrical variation of capacitance under ac excitation around a given de operating point. Back to back pairs of varactors also facilitate the dc biasing since either side of the varactors do not need to be dc blocked. Such an embodiment of a bandpass filter is shown in  FIG. 18  including five coupled resonators  10   j , each of which has associated with it a pair of back to back varactors  50   j ; this structure will cover more than 35 GHz to 70 GHz and typically has a 40 dB suppression at sub-harmonic frequencies, better than a 10 dB return loss, and insertion loss that has an amplitude equalization feature as shown in  FIG. 19 . That amplitude equalization feature is demonstrated in  FIG. 19  where y-axis represents insertion loss in dB, and the x-axis is the transfer function in pure numbers. Curves “0” to “14” represent the actual tuning voltage value in volts. As the tuning voltage is modified from 0 to 14 volts the center frequency of the response moves from around 32 GHz to somewhere around 65 GHz. From the y-axis one can tell that the insertion loss value changes from “high” loss (i.e. around 10 dB) to “low” loss (i.e. around 4 dB) as the frequency increases. This is the amplitude equalization feature. Equalization effect is due to the low reactance value of the resonators preferred for wide tuning bandwidth and the relatively high resistive components in the circuit such as the resistance of the varactors. As the filter is tuned higher in frequency, the reactance of the resonators increases while the overall resistance of the components stays relatively constant or decreases, the insertion loss of the circuit improves and amplitude equalization is achieved. Further control of the amplitude equalization slope can be achieved by introduction of additional resistance on the coupled lines as shown in  FIG. 20  where the additional resistances  120  are interconnected between each resonator cavity and ground. Bumping of pads and grounds, including control lines, by the use of bumps  130 ,  FIG. 21 , is implementable for flip chip mounting of the devices. The fully integrated nature and the availability of solid ground almost everywhere in the MMIC bandpass filter construction allows for the best of band rejection and ultimate in high frequency transition performance. 
     Although specific features of the invention are shown in some drawings and not in others, this is for convenience only as each feature may be combined with any or all of the other features in accordance with the invention. The words “including”, “comprising”, “having”, and “with” as used herein are to be interpreted broadly and comprehensively and are not limited to any physical interconnection. Moreover, any embodiments disclosed in the subject application are not to be taken as the only possible embodiments. 
     In addition, any amendment presented during the prosecution of the patent application for this patent is not a disclaimer of any claim element presented in the application as filed: those skilled in the art cannot reasonably be expected to draft a claim that would literally encompass all possible equivalents, many equivalents will be unforeseeable at the time of the amendment and are beyond a fair interpretation of what is to be surrendered (if anything), the rationale underlying the amendment may bear no more than a tangential relation to many equivalents, and/or there are many other reasons the applicant can not be expected to describe certain insubstantial substitutes for any claim element amended. 
     Other embodiments will occur to those skilled in the art and are within the following claims.