Abstract:
A method and apparatus are provided for modulating a laser beam with an information signal. The method includes the steps of modulating a plurality of coherent, optical signals with a respective information signal of the plurality of information signals and directing the modulated optical signals onto a photodiode. The method further includes the steps of detecting the collected signals in the photodiode and refractively synchronizing the laser beam with clock reference signals which allow easy separation of the modulated signal blocks at the receiving end of the fiber optic.

Description:
This application is a Continuation-in-Part of Ser. No. 08/811,207 filed Mar. 5, 1997 now U.S. Pat. No. 5,818,857. 
    
    
     FIELD OF THE INVENTION 
     The field of the invention relates to lasers and in particular to the stabilization and optical modulation of lasers used for cable television transmissions. 
     BACKGROUND OF THE INVENTION 
     The use of lasers for the transmission of information in communication systems is known. Such use has typically been limited to amplitude or phase modulated systems that in use often attain a speed of several megabytes. 
     Laser systems in current use for communications (e.g., solid state pumped 1550 nm lasers with 150 mW output available for fiber optic communication system from Laser Power Corp., San Diego, Calif.) are typically amplitude modulated because of certain inherent limitations in a laser&#39;s ability to change frequency. Lasers, in fact, are often limited to a single frequency or a narrow range of frequencies. The tendency of a laser to operate within narrow ranges is inherent in the resonant cavity used in the generation of laser signals. 
     A resonant cavity of a laser is designed to amplify optical signals of a desired frequency and attenuate signals of an undesired frequency. The cavity amplifies desired frequencies through use of a laser cavity dimensioned in one-quarter wavelength increments. The closer the cavity dimensions are to a desired tolerance, the narrower the range of frequencies within which the laser will operate (the linewidth). The narrower the linewidth, the less inherent amplitude and/or phase noise will be transmitted in a laser signal. Further, the narrower the linewidth, the more power is focused into a desired center frequency. 
     While gas lasers have been developed with extremely narrow linewidths, solid state lasers do not perform nearly as well. Distributed feedback (DFB) semiconductor lasers, in fact, are known to have relatively wide linewidths. 
     As the junction current of a DFB laser is changed (or the cavity temperature changes), the operating frequency of the laser also changes. Thus, the linewidth may also vary. Static variations in the inside cavity dimensions may cause the cavity to inherently resonate at a number of frequencies. Variations in the junction current may cause a center frequency to shift (i.e., hop) from one resonant regime to another. Changes in cavity dimensions caused by temperature changes will have a similar effect. 
     Because of their inherent low cost and reliability, DFB lasers have an enormous potential in laser communication systems. Consequently, a need exists for a method of controlling the linewidth of DFB lasers and for efficiently modulating such devices. 
     SUMMARY 
     A method and apparatus are provided for optically modulating a laser beam with an information signal. The method includes the steps of modulating a plurality of coherent, optical signals with a respective information signal of the plurality of information signals and directing the modulated optical signals onto a photodiode. The method further includes the steps of detecting the collected signals in the photodiode and refractively synchronizing the laser beam with the detected signals. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a block diagram of a stabilized laser transmission system in accordance with an embodiment of the invention; 
     FIG. 2 is a schematic view of the monochrometer of the system of FIG. 1; 
     FIG. 3 is a cut-away side view of the optoelectronic clock of FIG. 1; 
     FIG. 4 is a block diagram of the phase lock loop of FIG. 1; 
     FIG. 5 depicts a block diagram of an alternate optical modulator embodiment of the system of FIG. 1; 
     FIG. 6 depicts a block diagram of an electronic modulator version of the system of FIG. 1, under an alternate illustrated embodiment; 
     FIG. 7 depicts a block diagram of an electronic demodulator for use with the system of FIG. 1; 
     FIG. 8 depicts a block diagram of a sampling clock frequency filter of the system of FIG. 7; and 
     FIG. 9 depicts apparatus for generating even-powered, higher multiples of a base band clock frequency for use with a modulator of FIG.  1 . 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 1 is a block diagram of a laser optical modulation system  10  generally in accordance with an embodiment of the invention. Included within the optical modulation system  10  is a transmitting laser  12 , a first laser stabilization feedback section (circuit)  13 , a second laser stabilization feedback section (circuit)  14  and a laser optical modulation section  16 . 
     The transmitting laser  12  may be any distributed feedback (DFB) IR laser (e.g., 1.31 μm or 1.55 μm wavelength) compatible with an appropriate temperature control. The junction current of the laser  12  is provided by a current controlled source (not shown) supplying a rated current with no more that 0.1% ripple. 
     Temperature control of the laser  12  is accomplished using an active heat control device  18  and balancing heat source  20 . The active heat control device  18  can be a temperature control device sometimes called a thermo-electric control or TEC located within a mounting surface (e.g., a heat sink) of the laser  12 . The device  18  may be implemented using a thermocouple sensor and controller coupled to any thermally active temperature control device (e.g., a Peltier effect thermo-electric controller or heater/cooler). The operating temperature of the heat control  18  of the laser  12  may be held to an appropriate setpoint temperature (e.g., 32° F.) with appropriate limitation on temperature variation (e.g., no more than 0.1° F.). 
     The balancing heat source  20  is directed to stabilization of cavity dimensions by temperature control and may be implemented using an appropriately sized nichrome wire wrapped around the outside of the cavity of the laser  12  and surrounded by a thermally conductive, electrically non-conductive material (e.g., soreison, etc). The heat source  20  may be used to provide an appropriately stable cavity temperature (e.g., 32.1° F.±0.1° F.) to restrict cavity mode hopping of an output of the DFB laser  12 . 
     The balancing heat source  20  of the first feedback circuit  13  operates in conjunction with the second feedback circuit  14  to stabilize a natural emission and node frequency of the cavity of the transmitting laser  12 . While the first feedback circuit  13  uses the transmitting laser  12  as a reference, the second feedback circuit  14  compares an output of the transmitting laser  12  with an output of a paradigm laser  40 . 
     The paradigm laser  40  may be a low power reference diode laser operating with a junction current fixed to within 0.1% and operating at 1.31 μm or 1.55 μm wavelength. The paradigm laser  40  may be calibrated using an appropriate instrument standard (e.g., a Zeiss DK-2 Spectrophotometer using a quartz-iodine lamp that is NBS traceable) to a known energy level in each wavelength. The paradigm laser  40  may also be stabilized using a temperature controlled heat sink and an active temperature controller similar to that used by the transmitting laser  12 . 
     The fundamental problem associated with the stability of the transmitting laser  12  and paradigm laser has been determined to be control of the resonant modes operating within the cavity. Control of the resonant modes, in turn, is highly dependent upon the dimension of the laser cavity. The temperature of the cavity has been determined to be a significant factor in the cavity dimension and laser stability. Further, where attempts are made to control the cavity temperature, the cavity temperature often overshoots a set point due to the thermal lag (and thermal mass) associated with each laser  12 ,  40 . The solution to the problem, in fact, has been found to lie in control of the cavity temperature by modeling the laser cavity as a transient thermodynamic system. Using embedded thermistors and a summing operational amplifier, it has been found that the active temperature controller  18  can be adapted to follow the transient temperature using techniques previously described by the inventor (see for example, Palmer, J. R.,  Transient Heat Transfer in Flat Plates,  Vol. II Constant Temperature, Pro Se Publications, San Diego, Calif. (1995)). 
     In the case of a thermal system which, for the first surface, has characteristics of: 
     
       
           R   α &lt;(5{square root over (ατ)}+ R   o ),  t   o   &lt;6{square root over (ατ)} and    R   o &lt;6{square root over (ατ)}, 
       
     
     the convoluted transform will look like:            Δ                   T   1       =     Δ                   T   0            Ψ   o          (       h   c            Δ                   T   0            ∇       Ψ   2.75              α     ν             Γ       K   3.75                    -     β   1.75       +     μ   2.75     +     ζ   3.75           )           ,                          
     and the deconvoluted transform will have the form:            Δ                   T   1       =       (         2        (         Δ   0               (       τ                   R   0   2        π                   h   c         ρ                   c   p          R   α   2        π                 t       )         -     T   1       )          ατ        ρ                   c   p          R   α   2        π                 t       K                 τ                   R   0   2        π              Ψ   o          (       (     erfc        (   x   )       )     -     (     erfc        (     x   1     )       )     +     (     erfc        (     x   2     )       )       )         )     -     (         2        (         Δ   0               (       τ                   R   0   2        π                   h   c         ρ                   c   p          R   α   2        π                 t       )         -     T   1       )          ατ        ρ                   c   p          R   α   2        π                 t       K                 τ                   R   0   2        π              Ψ   o          (       (     erfc        (   y   )       )     -     (     erfc        (     y   1     )       )     +     (     erfc        (     y   2     )       )       )            x        (       erfc        (   y   )       -            (         t   o          H   o       +       H   o   2        ατ       )            erfc        (   z   )           )         )     +     (         2        (         Δ   0               (       τ                   R   0   2        π                   h   c         ρ                   c   p          R   α   2        π                 t       )         -     T   1       )          ατ        ρ                   c   p          R   α   2        π                 t       K                 τ                   R   0   2        π              Ψ   o          (       (     erfc        (   y   )       )     -     (     erfc        (     y   1     )       )     +     (     erfc        (     y   2     )       )       )         )     +     (         2        (         Δ   0               (       τ                   R   0   2        π                   h   c         ρ                   c   p          R   α   2        π                 t       )         -     T   1       )          ατ        ρ                   c   p          R   α   2        π                 t       K                 τ                   R   0   2        π              Ψ   o          (       (     erfc        (   x   )       )     -     (     erfc        (     x   1     )       )     +     (     erfc        (     x   2     )       )       )            x        (       erfc        (   y   )       -            (         t   o          H   o       +       H   o   2        α                 τ       )            erfc        (   z   )           )         )         ,                          
     where:            H   o     =       0.5          π   ατ         =     &gt;       t   0          cm     -   1               ,                z   =         t   o       2        ατ         +       H   o          ατ           ,                  x   1     =       R   o       2        ατ           ,                  x   2     =       R   α       2        ατ           ,                  ξ                 x     =     1.0     π         ,                          
     and: 
     ΔT°=T 1 -T 2 , in °C., 
     T 1 =constant source temperature in °C., 
     t 1 =starting temperature of the flat plate at time τ=0 in °C., 
     h c =heat transfer coefficient from temperature source in W/cm 2 °C., 
     F o-ang critical =absorbed flux density in W/cm 2 , 
     α=thermal diffuseivity of material in cm 2 /sec, 
     τ=time interval that the heat source is running in seconds, 
     K=thermal conductivity of the material in W/cm 2 °C., 
     t o =thickness of the flat plate component in cm, 
     c p =specific heat of the plate material in W-sec/g°C., 
     p=density of the plate material in g/cm 3 , 
     R o =radius of the heat source beam on the plate per cm, 
     R α =radius of the flat plate in cm for any thickness greater than t=0 cm. 
     The temperature rise at some depth into the component at t&gt;0 will follow from 
     
       
           R   α &lt;5{square root over (ατ)} R   o   , t   o   &lt;6{square root over (ατ)}, and    R   o &lt;6{square root over (ατ)}. 
       
     
     The convoluted transform will have the form:            Δ                   T   t       =       Δ                   T   0            Ψ   o          (       h   c            Δ                   T   0          ∇       ψ   3.75             Γ       K   4.75             )         +     Δ                   T   0            Ψ   o          (       h   c            Δ                   T   0          ∇       ψ   5.85             Γ       K   6.85             )             ,                          
     and when deconvoluted will have the form:          Δ                   T   t       =       (         2        (         Δ   0               (       τ                   R   0   2        π                   h   c         ρ                   c   p          R   α   2        π                 t       )         -     T   1       )          ατ        ρ                   c   p          R   α   2        π                 t       K                 τ                   R   0   2        π              Ψ   o          (       (     erfc        (     y   o     )       )     -     (     erfc        (     y   o1     )       )     +     (     erfc        (     y   o2     )       )       )         )     -     (         2        (         Δ   0               (       τ                   R   0   2        π                   h   c         ρ                   c   p          R   α   2        π                 t       )         -     T   1       )          ατ        ρ                   c   p          R   α   2        π                 t       K                 τ                   R   0   2        π              Ψ   o          (       (     erfc        (   x   )       )     -     (     erfc        (     x   1     )       )     +     (     erfc        (     x   2     )       )       )            x        (       erfc        (   y   )       -            (         t   o          H   o       +       H   o   2        α                 τ       )            erfc        (   z   )           )         )                              
     where:            y   o     =     t     2          α                 τ             ,                  y   o1     =           t   2     +     R   o   2           2        ατ           ,                  y   o2     =           t   2     +     R   α   2           2        ατ           ,                          
     and 0&lt;t≦t o . 
     As demonstrated, the active temperature controller  20  can be provided which follows the non-linear response of the thermal mass of the heat sink of the laser in real time. Using the equations demonstrated above, appropriate scaling and response factors may be provided which track the thermal characteristics of the transmission laser  12  and paradigm laser  40 . 
     Another factor is the control of the temperature of the laser cavity using the balancing heat source  20 . Control of the balancing heat source  20  is established using a first feedback loop  13  and a second feedback loop  14 . The first feedback loop  13  operates by stabilizing a natural emission frequency of the atoms of the cavity with a mode frequency of the cavity. 
     The objective of the first feedback loop  13  is to ensure that only a single laser frequency (mode)is present and that the laser  12  is stable in producing that single mode during operation. In general, the gain region of a DFB laser must be regarded as relatively wide. As such, several longitudinal modes of operation may normally be present during operation of a DFB laser. Under the invention, the gain region of the DFB laser  12  may be narrowed by adjusting the corrugations, but temperature control is still necessary. This is because the position of the gain profile depends on the bandgap and this phenomenon is extremely dependent upon variations in temperature. Consequently, a slight change in temperature could cause the position of the gain profile to shift sufficiently to cause the laser  12  to mode “hop” to another longitudinal mode more favorably disposed with regard to the gain profile. 
     As a means of developing a solution to the gain profile problem, it can be assumed that a single mode is oscillating at frequency v m  which is greater than the natural emission frequency of the atom, v o . The natural frequency and oscillating frequency v m  creates two “holes” in the distribution. The oscillation at frequency v m  is a standing wave within the cavity, consisting of two waves traveling in opposite directions. The two waves can be designated by “+” and “−”, respectively. Both waves have a frequency of v m . The interaction of the waves traveling in the positive +vm direction with the laser medium will be greatest for those atoms that have a velocity component direction of +v x , such that          v   m     =         v   0          (       1   +     v   m       c     )       .                            
     There are, therefore, two groups of atoms whose stimulated emission contributes to the laser output intensity. The population inversion is reduced for these atoms and gain saturation occurs with regard to these atoms. Two “holes are burned” and they are symmetrical about v x =0 and correspond to atoms with velocities of plus and minus v x . Under the embodiment, we can change the frequency of the oscillating mode until the peak frequency of the laser line is equal to the natural emission frequency (i.e., v m =v 0 ). This can be accomplished by varying the length of the cavity by small variations in the cavity temperature. In changing the temperature ever so slightly, only a single group of atoms can contribute to the lasing process, viz., those with zero-x component of velocity, and providing only a single ‘hole’ in the population inversion-velocity curve. When this happens, the output power has been found to drop as the available inverted population is smaller than before. 
     The small increase in power resulting from the slight deviation from the center frequency of the laser  12  is used as a feedback in the first feedback loop  12  to stabilize the frequency of the laser  12  at the line center by minimizing the output. Because of the power differences, any drift in the mode frequency will cause one mode to increase in power and the other to decrease. The first feedback loop  13  is used to monitor the two different power levels and to subsequently to provide a feedback signal to control the cavity length. 
     Fundamentally, it has been found that adjacent cavity modes are plane polarized with their planes of polarization at right angles to each other. In order to isolate and capture the two cavity modes, the output signal of the laser  12  is split into two beams in beam splitters  24 ,  26 . A polarizer  28 ,  30  inserted into the path of each beam. Each polarizer    28   ,    30    is oriented orthogonally with respect to the other, to isolate the two adjacent cavity modes. Two matched InGaAs detectors are used to detect the two modes. The difference in the output of the detectors  28 ,  30  is proportional to the difference in power of the two modes. 
     The output of the two detectors    28   ,    30    is provided as an input to a differential input voltage controlled oscillator (VCO)  32 . The differential input from the two detectors  28 ,  30  is used by the differential input VCO  32  to adjust the power to the heating coil  20  to, in turn, change the length of the optical cavity of the laser  12 . This, in turn, alters the mode frequencies of the laser  12 . The operation of the first feedback loop  13  has been found to stabilize the frequency of the laser  12  to within 10 Å line width. 
     To further stabilize the laser  12 , a second feedback loop  14  is provided. Within the second feedback loop  14 , an output beam of the laser  12  is compared with an output beam of a reference paradigm laser  40  and the difference provided as a second feedback signal as a further means of achieving control over the stability of the laser  12 . 
     Using a beam splitter  26 , a small portion of the output beam of the laser  12  is provided as a first input to an optoelectronic clock  34  through a first monochrometer  34 . A second input to the optoelectronic clock  34  is provided from the paradigm laser  40 , through a second monochrometer  38 . Under the embodiment, the first and second monochrometers  34 ,  38  may be constructed substantially the same. 
     Within the monochrometer, the beam goes through a slit  50  (FIG.  2 ), passes through a beam expander-collimator  52  and then on to a grating  54 . From the grating  54 , the beam goes to a parabolic mirror  56  with a 24 cm focal length. 
     The slit  50  is fabricated with characteristics which limit the bandwidth of incident radiation. Under the embodiment, the dimensions of the slit  50  may be determined using an appropriate bandwidth limiting technique (e.g., see Richardson, D.,  Diffraction Gratings,  Applied Optics and Optical Engineering, Vol. V, Part II, Optical Instruments, R. Kingslake, ed., Academic Press, New York, N.Y., (1969)). 
     The beam expander and collimator  52  are of conventional design for a 1.31 μm signal, using quartz optics. The grating  54  may be fabricated with a blaze angle of 24° and a line spacing of 13.4(10 3 )lines/inch or 527.559 lines/mm. 
     Where light is incident at an angle, α, relative to the surface, normal to a reflecting grating, ruled with a spacing, d, the path difference for the light incident on any two adjacent grooves is dsin(a). When this light is diffracted from the rulings at some other angle, β, the path difference for the light is further increased by the amount of dsin(β). The reflected light of wavelength, λ, will be in phase over the entire wave front when the path difference for rays incident on adjacent grooves is an integral multiple of the wavelength. For light of a given wavelength incident at a particular angle, the light reflected from all of the grooves will be in phase only at certain angles. The number of wavelengths of a path difference from adjacent grooves is called the order of interference, mλ. Using the described variables, a grating equation can be written as follows: 
     
       
           mλ=d (sin α±sin β). 
       
     
     When P is equal to the blaze angle θ, then λ β , the first order blaze wavelength can be described as: 
     
       
         λ β =2 d  sin (θ). 
       
     
     For a grating  54  where θ is 24° and the grating spacing is 527.559 lines/mm, the spacing, d, is equal to 1.89553 μm. From the equation, λ β  can be determined as follows: 
     
       
         λ 60 =2  d  sin (θ)=2(1.89553 μm) (sin 24°)=1.541956 μm. 
       
     
     If the blazed wavelength, λ β  for the condition α=β is known, and the blazed wavelength λ β ′ for other combinations of α and β is desired, then the desired values may be determined as follows:          λ   β   ′     =       λ   β          cos        (         α   ′     ±   β     2     )                                
     For the particular case of λ′ β =1.31 μm, where we want to find the angle normal to the grating that is suitable for our wavelength, the angle may be determined by:              λ   β   ′       λ   β       =     cos        (         α   ′     ±   β     2     )         ,                          
     which may be simplified to produce;            1.31   1.541956     =     0.843085   =     cos        (         α   ′     ±   β     2     )           ,                          
     and 
     
       
         cos −1 (0.84305)=32.53266°. 
       
     
     From this value, α′ may be determined as follows: 
     
       
         α′=2(32.53266)+24=89.9653°. 
       
     
     Another issue regarding the grating  54  is that of resolution. Using a 6 inch wide grating, the grating resolution can be determined using Fraunhofer diffraction theory from the following expression:          λ   Δλ     =       Nd        (       sin                 α     -     sin                 β       )       λ                            
     where:              N   =     number                 of                 grooves                 =     13.4                   (     10   3     )                     lines   /   inch     ×   6                 inches                   =     80.4                   (     10   3     )                   lines       ,                                
     and              d   =                spacing                 between                 grooves                   (   μ   )                   =                1.89552                   (     10     -   6       )                   meter                 =                1.89552                   μm   .                                    
     Substituting N and d into the equation results in the expression as follows:          λ   Δλ     =                   80.4        (     10   3     )                   lines   *   1.89552        (     10     -   6       )                   meters   *                     (     0.9998669   -   0.4067366     )             1.30        (     10     -   6       )                   meters                 λ   Δλ     =       9.039294        (     10     -   2       )                   meters       1.30        (     10     -   6       )                   meters                     Δλ   =         (     1.31        (     10     -   6       )                   meters     )     2       9.039294        (     10     -   2       )                   meters                   =     0.189848        (     10     -   10       )                   meters                 =     0.189848                 Angstroms                                  
     The resolving power of a grating is a measure of its ability to separate adjacent spectrum lines. It is expressed as λ/Δλ, where λ+Δλ is the wavelength of a spectrum line that is just barely distinguishable from a line at wavelength λ. 
     Using the techniques developed herein, it should be evident that large gratings used at high angles are needed to achieve high resolving power. The actual attainment of high resolving power with a grating depends upon the optical quality of the grating surface, the uniformity of the spacing of the grooves, and the associated optical components. The equation suggests that one should be able to have an infinite resolving power simply by increasing the total number of grooves in a given width. However, there is a fundamental rule that has to be applied, (i.e., that it is absolutely necessary that the incident wavelength be less than the groove spacing (d)). If not for the fundamental rule, a grating having 4500 lines/mm would have been selected. Unfortunately, this would have provided a groove spacing of 0.222 μm, which is 5.85 times too small to provide an effective solution to the problem. Using the techniques described herein, the resolution of the laser  12  has been improved to provide a line width that is approximately 0.18696 Å 1.8696(10 −11 ) meter. That is, the line width of the grating  54  spans the range of between 1.300018 and 1.299981 μm. 
     Effectively, the line width is stable to 5 parts in 10 11 . The approximate bandwidth can be shown as follows:          ΔΦ                 L     =         3.0        (     10   8     )         4        (     Δλ   λ     )          Y   m         =     2.6075        (     10   15     )                                
     where Y m =0.002, which is a dimensionless material dispersion coefficient. 
     As mentioned above, the paradigm laser  40  is a low power diode laser operating at 1.31 μm wavelength. Prior to entry into the paradigm monochrometer  38 , the laser beam from the paradigm laser  40  is passed through a 300 Å line filter as a rough control before entry into the monochrometer  38 . 
     Within the monochrometer  38 , the beam passes through the expander-collimator  52 , onto the grating  54 , and is reflected from the parabolic mirror  56 , as described above and is passed on by the rotating mirrors  62  and grating  66  of the optoelectronic clock  34 . After passing through the grating  66 , the beam strikes a InGaAs detector  70  matched to another InGaAs detector  68  detector on the other side of the optical phase lock loop  36 . As mentioned above, the paradigm laser  40  is calibrated using the Zeiss DK-2 Spectrophotometer to precisely determine the optical output of the laser  40 . 
     With the transmitter laser  12  slaved to the paradigm laser  40  through the optical phase lock loop  36 , the variable controlled output  54  will feed back to the cavity temperature control  20  of the transmitter laser  12  as small increments of voltage control on the heating element surrounding the optical cavity of the transmitting laser  12 . In this manner, it has been found that temperature (and length) of the cavity can be more precisely controlled. The feedback signal  54  of the second feedback circuit  14 , in fact, has been found to provide a granularity of control 53 times better than the first feedback circuit  13 . 
     Turning now to the optoelectronic clock  34 , the optoelectronic clock  34  forms a portion of the second feedback circuit  14 . For purposes of simplicity, FIG. 3 will be used to explain this function. 
     The essence of the clock  34  is two counter-rotating disks  72 ,  74 . The two disks  72 ,  74  may be driven by variable speed motors (e.g., Dremel motors). The variable speed capability of the disks  72 ,  74  is such that the speed may be continuously varied from a low of 83.33 revolutions per second (rps) to 500 rps. Under an embodiment, the speed of the inner disk  72  is chosen to be 500 rps. The two disks  72 ,  74  rotate in opposite directions. 
     A number of mirrors  60 ,  62  are mounted to the inner mirror disk  72 . More specifically, sixteen mirrors  60 ,  62  are mounted at a 45° angle to a rotational axis of the disk  72  around the periphery of the disk  72  at a radius of 63.5 mm. The mirrors  60 ,  62  are mounted such that a light beam traveling parallel to the axis of rotation of the disk  72  would be reflected radially through a rotating transmissive grating  64 ,  66  on the second disk  74  at a predetermined angle of the disks  72 ,  74  to strike a set of stationary detectors  68 ,  70 . 
     The second disk  74  rotates at a speed of 500 rps. The second disk may be equipped with an annular ring  76  of a radius of 97.0209 mm, fitted with a rotating transmissive grating  64 ,  66 . 
     The total cycles per second produced by the optoelectronic clock  34  may be determined by the equation; 
     
       
           v   o =αβ( N   m   X+Y ), 
       
     
     where: 
     α=grating lines/mm, 
     β=circumference of the second disk, 
     X=rps of the mirror disk, 
     Y=rps of the second disk and 
     N m =number of mirrors on the mirror disk. 
     For X and Y of 500 rps with 16 mirrors on the inner rotating shaft and a second disk of circumference 609.60029 mm with a grating density of 905.5 lines/mm, the output frequency v o  is 4,969,194,103 Hz or 4.969 GHz. As should be apparent from the equation, the frequency of the signal detected by the detectors  68 ,  70  can be adjusted by changing the gratings, the number of mirrors, the radius of the mirrors, or the radius of the grating. Having passed through the grating  64 ,  66 , the laser beam strikes the detectors  68 ,  70  providing an electrical signal that is conditioned and amplified to establish a particular tuning frequency. 
     For purposes of the second feedback circuit  14  alone, the speed of the counter-rotating disks  72 ,  74  is not considered critical. As a consequence, the disks  72 ,  74  may be operated at a speed convenient for generation of reference frequencies. Of more importance for purposes of the second feedback circuit  14  is the ability of the disks  72 ,  74  to chop a light beam from both the transmitting laser  12  and from the paradigm laser  40  into light pulses that may be compared as to phase and intensity. 
     FIG. 4 is a block diagram of the phase lock loop circuit  36 . 
     As shown, an output of the sensors  68 ,  70  is shaped within an op amp  102 ,  104  and Schmidt trigger  106 ,  108 . The shaped pulses are passed through a loop filter  130 ,  132  before comparison in a control logic section  128 . 
     From the detectors  68 ,  70 , the level of the signal requires amplification to raise the signal to a sufficient level to trigger the Schmidt trigger  106 ,  108 . The signal from the detector  70  of the paradigm laser  40  uses a non-inverting op amp  104 . The signal from the detector  68  of the transmitting laser  12  may use an inverting op amp  102 . It should be noted that it is not all that important that the input signal from the detectors  68 ,  70  match, but only the square wave of the Schmidt triggers  106 ,  108  substantially match. 
     The loop filters  130 ,  132  provide a means of averaging an output of the phase lock loop  36 . A signal from the loop filter  130 ,  132  is compared with an output of the Schmidt trigger  106 ,  108  in control logic  128  (e.g., an XOR gate) to identify and reinforce coincidence of phase and amplitude relationships between the transmitting laser  12  and paradigm laser  40 . The output of the control logic  128  is then scaled and provided as a control output of the second feedback circuit  14  through the heater control logic  20 . 
     FIG. 5 is a block diagram of an illustrated example 150 of the laser modulation section  16  of the system of FIG.  1 . While the modulator  44  of FIG. 1 is shown functionally in the form of a mirror which modulates a phase of an incident light beam based upon movement of the mirror&#39;s surface, it is to be understood that the laser modulator section  16  may be implemented under any of a number of formats (e.g., the systems shown in FIGS.  4 - 5 ). 
     Similarly, the optoelectronic clock  45  may also be implemented under a number of formats. Under the embodiment, a predetermined number of information channels (e.g., 140 television signals) of an appropriate bandwidth (e.g., 6.2 MHZ each) may be grouped and be used to modulate an optical subchannel (e.g., of a 1 GHz bandwidth). A number of subchannels, in turn, may be combined and transmitted over the fiber  50 . Each optical subchannel may be generated through the use of its own clock reference  45  which clock reference signal may be included as a portion of the information on the subchannel and recovered as a timing reference at a receiver. 
     The optoelectronic clock  45  may be implemented using a suitable high frequency reference (e.g., a quartz crystal) and a number of frequency multipliers to generate a multitude of reference clock frequencies from a base frequency. For example, phase-locked loop (PLLs) frequency multipliers are known which use integer dividers in a feedback path and a voltage controller oscillator (VCO) in a feedforward path to generate stable multiples of a base clock frequency. Alternatively, fractional frequency dividers using sigma-delta modulation of the feedback divider may be used for the generation of sub-integer multiples of the base frequency. 
     Under the illustrated embodiment of the modulator  150  of FIG. 5, a number of optical subchannels  1 -n (two channels  152 ,  154  shown in FIG. 5) may be combined within a combiner  156 . An output signal of the combiner  156  may be used to modulate a transmission laser  158 . Each optical subchannel  152 ,  154  in turn may be used to encode a block of information signals from a signal provider  160 ,  162 . 
     Where the signal provider  160 ,  162  is a television signal source, the signal from the provider  160 ,  162  may include up to 140 channels, with a bandwidth per channel of 6.2 MHZ. Encoding of the channels into a spectral information block by the signal provider  160 ,  162  may be accomplished under any of a number of formats (e.g., frequency division multiplexing (FDM)). Where FDM is used, the first channel may occupy a center frequency of 3.1 MHZ. A 1 MHZ guardband may be provided between an upper frequency limit of the first channel and a lower frequency limit of the second channel (i.e., between 6.2 and 7.2 MHZ). The second channel may occupy a center frequency of 10.3 MHZ. The third channel may occupy a center frequency of 17.5 MHZ, and so on. Including guardbands, the modulated signal from the provider  160 ,  162  may have a total bandwidth of approximately 1 GHz. 
     Modulation of the signal from a signal provider  160 ,  162  onto an optical subchannel  152 ,  154  may be accomplished under any of a number of methods. Under a first method, an electro-optical differential multiplexer  168  may be used to encode (e.g., refractively synchronize) the signal from the signal provider  160  onto a laser signal from a first laser source  166  of the first optical subchannel  152 . 
     The electro-optical differential multiplexer may be any electro-optical device capable of modulating an optical signal with a controlling electrical signal through refractive synchronization. For example lithium niobate (LiNb0 3 ) or gallium arsenide (GaAs) crystals are known to have such characteristics. Suitable devices for use with the modulator  150  include those devices with a suitable radio frequency (RF) interface and internal laser source or optical interface for an external laser source (e.g., a Marconi Model Lt 7000 with External Modulator, External Modulator by UTP SITU, Marconi LC 1000, etc.). 
     Optical refractive synchronization is a process whereby a controlling signal is used to modulate an optical signal. Fundamentally, the process of refraction is the change of energy, direction or speed of a light beam which is propagating through a medium. In a first case, the change in direction may be a continuous bending of the light beam and of the subsequent change of the speed of light in the medium which is referred to as the index of refraction of the material. In a second case, there is an abrupt change in the index, polarity, or phase of the medium which directs the energy out of the medium, or changes the ability of light to pass through the medium, thereby absorbing the energy. This second case is the technique employed in optical refractive synchronization. The light is passed through a crystal which has the refractive index changed abruptly, by imposition of an electric field, by passing a controlling signal through the crystal, which then causes the crystal to allow the light to pass through, or be absorbed in the crystal at the frequency of the controlling signal. This optical modulation is performed, therefore, through control of the refraction and polarization characteristics of the optical crystal. 
     By combining the frequencies of various sources and optically modulating them onto a respective clock frequency with an appropriate sampling frequency and amplitude ratio, it is possible to synchronize all of the information signals to their respective optical light beams. It then also becomes possible to combine them with other light beams with similar optical modulation at singularly different clock frequencies and to nest these optically modulated beams onto yet another yet higher clock frequency by using optical diffractive synchronization. 
     Once the signal from the laser  166  is modulated with the information signal by refractive synchronization, the signal may be sampled within a second electro-optical differential multiplexer  170  (e.g., an External Modulator by UTP SITU APE, Marconi LC 1000, etc.). Sampling may be accomplished at any clock frequency with an appropriate sampling rate as determined from the Palmer Sampling Modulation Transfer Function equation as follows:          f     0      sample       =       2   π          (         cos     -   1            (     A     A   0       )       -     (       A     A   0              1.0   -       (     A     A   0       )     2           )       )                              
     where A is the modulated signal amplitude, Ao is a clock frequency amplitude. 
     For the case where a noise signal is included as part of the baseband signal, the equation can be modified as follows:          f     0      sample       =       2   π          (         cos     -   1            (     A     A   0       )       -       A     A   0              1.0   -       (     A     A   0       )     2             )          (       (       2          J   1          (   M   )         M     )          (     1.0   -     A     A   0         )       )                              
     where M is 8πδA/A o  and δ is baseband noise amplitude/2A o  and J 1 (M) is a Bessel Function of the First Kind of Order 1. Further, a required minimum clock sampling frequency for A 0 =(1/f 0sample )f max  where f max  is the highest frequency represented in the modulated signal A. From a practical point of view the best results have been found to occur where A/A o &lt;1.0 (i.e., the clock amplitude A o  is 2.0 to 7.0 times larger than the modulated signal amplitude A). 
     Further, since the sampling multiplexer  170  also functions to shift a subchannel operating frequency to a non-interfering spectral location for transmission through the optical cable  50 , the frequency of the square or sine wave clock  172  must be chosen with consideration given to adjacent subchannels (e.g., the second subchannel  154 ). 
     For example, where a bandwidth of the signal provider is in the area of 1 GHz, a frequency chosen for the first clock  172  may be 2.5 GHz. A frequency chosen for a second subchannel may be 3.6 GHz. A third subchannel (not shown) may be 4.7 GHz. Given that the channel capacity of the modulator  150  is believed to be 2200, or more, television signals (and using 140 television signals per subchannel) the number of subchannels  152 ,  154  that may be used in the modulator  150  may be up to, or exceed, 16. Clock frequencies that may be used by subchannels  4 - 16  may be as follows: 5.8 GHz, 6.9 GHz, 8.0 GHz, 9.1 GHz, 10.2 GHz, 11.3 GHz, 12.4 GHz 13.5 GHz, 14.6 GHz, 15.7 GHz, 16.8 GHz, 17.9 GHz and 19.0 GHz. However, these values are nominal and will be increased greatly by using 60 GHz or 100 GHz modulators. 
     When imposing signal modulations or carrier clock frequencies using a 12 GHz modulator or better, it is possible to synchronize two clock signals 180 degrees out of phase via selective time delays constructed with differing lengths of fiber optic. Such synchronization may be implemented, for instance, by adjusting a length of the optical fiber between the first and second clock modulators  402 ,  404  shown in FIG.  9 . Such method may be useful to speed up the reference clocks with only limited bandwidth on the clock sources and modulators. 
     The index of refraction of the 5 μm core of a fiber optic at 1.550 nm is assumed to be 1.458. Since this index of refraction represents the relative slowing of the velocity of propagation from that of free space, we can easily calculate the length of fiber optic necessary to induce a 180 degree delay at 12 GHz. The period of a 12 GHz wave is 1/(12.0( 10   9 )Hz)=83.33( 10-   12 ) sec. So the delay required is one half of 83.3 ps=41.667 ps. Since light travels at 299,776 km/s or approximately 3.0(10 10 ) cm/s, a delay of 41.667 ps represents about 8.5695 mm of fiber optic between modulators Since (3.0(10 10 ) cm/s/1.456)*(41.667 ps)=8.5696 mm. The fiber can be of length N*1.71392 cm+8.5696 mm where N is an arbitrary integer and still work correctly to multiply the two source signals together. There is nothing sacrosanct about 12 GHz. The same technique may be used for 60 or 100 GHz modulators, etc. 
     Under the technique of FIG. 9, a higher even-powered (2 n , n=1,2, . . . ) multiple of a base band clock frequency signal may be delivered to the final dual-ported information encoding modulator  404  from a lower frequency clock source  408  based upon the phase delay implemented by lengths of fiber optic cable  406  between (2 n −1) type  402  (single port) modulators in the chain of 2 n  modulators terminated by a single type  404  (dual ported) modulator. The phase delay implemented by the length of fiber between adjacent modulators is 2π/2 n  radians, where n is an integer. For example, the 5 GHz source  408  can provide a doubled 10 GHz (2 1  times 5 GHz) to the modulator  404  based upon a 180 degree (n radians=2π/2 1  radians) phase shift provided by the proper length of fiber  406 . If there were four modulators consisting of three type  402  and one type  404  (n=2, therefore a total of 2 2  modulators required) with the phase delay between adjacent modulators of 90 degrees (or n/2 radians=2π/2 2  radians), then a 20 GHz clock signal would be delivered to the last (dual-ported) modulator  404  from a 5 GHz base-band clock frequency source  408 . In general, any clock source  408  may be scaled upwards by a multiple of 2 n  using a total of 2 n  modulators 2 n −1 of type  402  and one of type  404  with n a positive integer, the last modulator  404  of which is dual ported to also allow input of the signal block. 
     With regard to FIG. 5, the multi-channel modulator  400  of FIG. 9 may be used in place of the subchannel modulator  152 . More to the point, a number of subchannels  152  (operating on different subcarrier frequencies) may be provided by the multi-channel modulator  400  where the only differences between modulators  400  are the number of modulators in the chain (divisible by 2 n ) and the length of the optical fibers  406  between them. 
     Under another illustrated embodiment, the subchannels  152 ,  154  may encode a signal from a signal provider  162  using direct current modulation. Under the embodiment, a current driver  174  is interposed between the signal provider  162  and DFB laser  176 . The current driver  174  modulates a junction current of the DFB laser  176  in a known manner to frequency modulate the optical signal with the information signal (e.g., 140 television signals). 
     To further improve the performance of the subchannels  152 ,  154 , the DFB lasers  166 ,  176  may be stabilized to reduce frequency drift due to environmental factors. Stabilization techniques that may be applied may include the temperature control systems described with reference to the stabilized laser  12  including modulators. More specifically, the techniques described in association with reference laser  12  may be advantageously applied to any laser application described herein, including modulators. 
     Following modulation of the subchannels  152 ,  154  by information signals from the signal providers  160 ,  162 , the information of the subchannels is combined. Combining may be accomplished by bringing the optical signals of the subchannels together and detecting the transmissions of each subchannel in a high speed photodiode  190 . 
     The step of bringing an optical signal of each subchannel  152 ,  154  together may be accomplished through the use of Cassegrain collection optics. To use the Cassegrain collection optic, an output end of an optical cable  196 ,  198  of each subchannel is collected and the body of each cable  196 ,  198  held in a parallel relationship with the bodies of the cables  196 ,  198  together forming a cylindrical shell  182  at an input to the Cassegrain collector  184 . The output end of each cable  196 ,  198  combines with the output ends of other cables of other subchannels to form an annular ring with an output of each cable directed into the collector  184 . 
     The signals from the subchannels  152 ,  154  strike a first and second parabolic mirrors, followed by a collimating and collection lens assembly  186 ,  188 . The combining of the optical signals occurs upon detection of the signals within the photodetector  190 . 
     The photodetector  190  may be any high-speed (e.g., 60 GHz) integrating photodiode. The photodiode, for example, may be a Newport Model PX-D7, or equivalent. 
     The signals combine in the photodiode  190  to form a modulation signal that, in turn, may be used to drive and frequency modulate a signal from a final transmission laser  192 . The transmission laser  192  may be any solid state laser, but in one preferred embodiment is a 1.550 μm DFB laser; another embodiment would be a 1310 nm laser. 
     The modulator  194  may be a LiNb0 3 , GaAs, or equivalent crystal and may function as an electro-optical diffraction multiplexer. Since the modulator  194  must operate at a higher speed (e.g., 100 GHz), it may be of a high speed variety (e.g., a Marconi LC 1000 Series Special, UTP SITU APE, etc.). 
     Under an alternate illustrated embodiment, the modulation section  16  of FIG. 1 may be implemented as shown in FIG. 6 by the modulation system  200 . Under the embodiment, modulation, mixing and frequency translation may be accomplished under a RF format. 
     As with the modulator  150  of FIG. 5, the modulation of source signals (e.g., 140 - 6.2 MHZ television signals) may be accomplished by first grouping the signals into 1 GHz wide information blocks and then frequency translating the 1 GHz information blocks into non-interfering positions over a transmission spectrum. 
     In the system  200  shown in FIG. 6, a grouping of signals (e.g., 140) is received as an input  216 ,  226  by a subchannel  202 ,  204 ,  206 ,  208 . Upon receipt of the signal, the signal is modulated by a clock signal from a voltage controlled oscillator (VCO)  224 ,  234 . The magnitudes of the clock signal and groups of signals may be controlled to ensure that the clock signal has a signal level as discussed above of 2-7 times higher than the information block. 
     As above, the clock frequencies of the VCOs  224 ,  234  may be chosen to place the blocks of information signals into non-interfering spectral locations of a final drive signal used for modulating the laser  158 . Since the blocks of information signals may have a bandwidth of 1 GHz, a clock frequency of the first VCO  224  may be chosen in the area of 2.5 GHz. The clock frequency of the second VCO  234  may be chosen in the area of 3.6 GHz. The clocks of the third and fourth subchannels may be chosen in the areas of 4.7 and 5.8 GHz, respectively. Subchannels  5 - 16  (not shown) may use similar frequency spacing. 
     Modulating the block of source signals by the clock signal results in sum and difference signals, as well as a strong signal component at the spectral location of the clock frequency. The modulated signal from the modulator(e.g.,  218 ) is bandpass filtered in a filter  220  to leave the difference and clock components of the modulation step. Amplification with an amplifier  222  may be necessary before combining the modulated signals of each subchannel  202 ,  204 ,  206 ,  208 . 
     Following amplification, the subchannel components may be combined in a set of mixers  210 ,  212 ,  214 . Since 16 inputs (one from each subchannel) to a single mixer may result in objectional artifacts, a two-step process is used. First pairs of channels  202  and  204 ,  206  and  208  are mixed in a first set of mixers  210 ,  212 . The final mixing step may occur in a final mixer  214 . 
     Modulation of the mixed signal from the mixer  214  may be applied as an input to the driving modulator  194  of the modulation system  158 . As described above, the signal may be applied to a electro-optical differential multiplexer (e.g., a 100 GHz LiNb0 3  or GaAs modulator) having an appropriate electro-optical response (e.g., a Marconi LC 1000 Series Special, UTP SITU APE, etc.). Similarly, the driver laser  192  exciting the modulator  194  of the transmission laser  158  of the modulation system  200  may be an infrared DFB laser of a 1.55 μm wavelength. 
     FIG. 7 depicts a demodulator  300  in accordance with an illustrated embodiment of the invention. Under the embodiment, a detector detects an optical signal from the fiber  50  and passes the detected signal to a divider  304 . Within the divider, a series of subchannel signals are separated from the transmitted signal based upon spectral location. A series of subchannel processors  306 ,  308 ,  310 ,  312  recover a clock signal and function to frequency shift the subchannel signal to a baseband frequency, thereby recovering the block of information signals originally transmitted. 
     Detection of the transmitted signal from the fiber  50  may be accomplished using a high speed photodiode  302 . An integrating sensor having a 60 GHz capability may be selected as being appropriate (e.g., a Newport PX-D7 diode). 
     The divider  304  functions to divide the signal detected in the detector  302  into a series of spectrally divided constituent parts. Functionally, the divider  304  acts as a multi-band passband filter where a different spectrum of the signal received from the detector  302  is passed to each of its outputs  338 ,  340 ,  342 ,  344 . For example, the first subchannel  306  may process a subchannel signal with a clock signal of 2.5 GHz and associated information signal. A second subchannel  308  may process a subchannel signal with a clock frequency of 3.6 GHz. 
     The passbands of each output  338 ,  340 ,  342 ,  344  of the divider  304  coupled to a respective subchannel  306 ,  308 ,  310 ,  312  may be determined based upon the relative location of the information signal with respect to the clock signal. For example, the first subchannel  306  may operate with a clock frequency of 2.5 GHz and a 1 GHz information signal (including a spectral block of 140 TV signals). When the 1 GHz information signal is encoded using the system  150  of FIG. 5, the 1 GHz signal is centered around the clock signal. In contrast, when the 1 GHz information signal is encoded using the RF process of the system  200  of FIG. 6, the information signal is offset to the spectral location of the difference component of the modulation operation. 
     Where the optical signal received by the receiver  300  was encoded by the encoder  150 , the passband of the first output  338  may be chosen as being in the area of 2.0 to 3.0 GHz. Similarly, the passband of the second output  340  of the divider may be chosen in the area of from 3.1 to 4.1 GHz. 
     Where the optical signal received by the receiver  300  was encoded by the encoder  200 , the passband of the first output  338  may be chosen as being in the area of 1.5 to 2.5 GHz. Similarly, the passband of the second output  340  of the divider may be chosen in the area of from 2.6 to 3.6 GHz. 
     Following separation of the subchannel components in the divider  304 , the subchannel signal may be amplified in an amplifier  314 ,  326 . Following amplification, the subchannel signal enters a splitter  316 ,  328  where a clock signal may be recovered from the subchannel signal. The splitter  316 ,  328  may include a pair of notch filters that passes a clock frequency to a sampling frequency filter  320 ,  332  and attenuates the clock signal from passing to the data bandpass filter  318 ,  330 . 
     A bandpass filter  318 ,  330  following the splitter  316 ,  328  also serves to further attenuate subchannel frequencies outside the proper spectral range. For example, the passband for the first subchannel may be chosen in the range of 1.5 to 2.5 GHz or 2.0 to 3.0 GHz, depending upon the transmitter  150 ,  200  used. The proper bandwidth of the bandpass filter  318 ,  332  for purposes of the receiver subchannels  306 ,  308  would substantially be the 1 GHz bandwidth of the information signals from the signal providers  160 ,  162 ,  216 ,  226  of the transmitters  150 ,  200 . 
     Within the sampling frequency filter  320 ,  332  a local voltage controlled oscillator (VCO)  348  (FIG. 8) may be used to duplicate a transmitter clock frequency. For instance, the local VCO  348  of the sampling frequency filter  320  of the first subchannel  306  may be configured to duplicate the clock  172  of the first subchannel  152  of the transmitter  150  or the VCO  224  of the first subchannel  202  of the transmitter  200 . Similarly, the local VCO  348  of the second subchannel  308  of the receiver  300  may be configured to duplicate the clock  180  of the second subchannel  154  of the transmitter  150  or the VCO  234  of the second subchannel  204  of the transmitter  200 . 
     A phase locked loop (PLL)  352  may also be used within the sampling frequency filter  320 ,  332 . The PLL  352  may be used to synchronize the local VCO  348  to the clock frequency recovered in the splitter  316 ,  328 . 
     Within the demodulator  322 ,  324 , the bandpass filtered, information signal may be translated to a baseband frequency by mixing the information signal from the bandpass filters  318 ,  330  with the recovered clock frequency from the sampling frequency filter  320 ,  332 . The translated baseband signal may then be low-pass filtered to remove any artifacts associated with demodulation and to recover the originally encoded information signal. An output driver  324 ,  336  may be used to transfer the recovered information signal to a channel processor for separation of each of the originally encoded (e.g., 140 television) channels. 
     A specific embodiment of a novel method and apparatus for construction of a stabilized laser according to the present invention has been described for the purpose of illustrating the manner in which the invention is made and used. It should be understood that the implementation of other variations and modifications of the invention and its various aspects will be apparent to one skilled in the art, and that the invention is not limited by the specific embodiments described. Therefore, it is contemplated to cover the present invention any and all modifications, variations, or equivalents that fall within the true spirit and scope of the basic underlying principles disclosed and claimed herein.