Abstract:
A method and apparatus for demodulation in high rate CDMA wireless communication is described. In a described high rate CDMA wireless system, a transmitter forms a set of individually gain adjusted subscriber channels using a set of orthogonal subchannel codes having a small number of PN spreading chips per orthogonal waveform period. An illustrative high rate CDMA wireless system uses Walsh codes, each having a duration of fewer than sixty-four chips per orthogonal waveform period. A receiver demodulates each of the subscriber channels using the same orthogonal subchannel codes.

Description:
This application is a divisional of application Ser. No. 08/856,428, filed May 14, 1997, which is a continuation in part of application Ser. No. 08/660,438, entitled “REDUCED PEAK-TO-AVERAGE TRANSMIT POWER HIGH DATA RATE CDMA WIRELESS COMMUNICATION SYSTEM” filed Jun. 7, 1996, now U.S. Pat. No. 5,926,500, issued Jul. 20, 1999 to Odenwalder, and a continuation in part of application Ser. No. 08/654,443 entitled “HIGH DATA RATE CDMA WIRELESS COMMUNICATION SYSTEM” filed May 28, 1996, now U.S. Pat. No. 5,930,230, all assigned to the assignee of the present invention. 
    
    
     BACKGROUND 
     I. Field 
     The present invention relates to communications. More particularly, the present invention relates to a novel and improved method and apparatus for high data rate CDMA wireless communication. 
     II. Description of the Related Art 
     Wireless communication systems including cellular, satellite and point-to-point communication systems use a wireless link comprised of a modulated radio frequency (RF) signal to transmit data between two systems. The use of a wireless link is desirable for a variety of reasons including increased mobility and reduced infrastructure requirements when compared to wire line communication systems. One drawback of using a wireless link is the limited amount of communication capacity that results from the limited amount of available RF bandwidth. This limited communication capacity is in contrast to wire-based communication systems where additional capacity can be added by installing additional wire line connections. 
     Recognizing the limited nature of RF bandwidth, various signal processing techniques have been developed for increasing the efficiency with which wireless communication systems utilize the available RF bandwidth. One widely accepted example of such a bandwidth-efficient signal processing technique is the IS-95 over the air interface standard and its derivatives such as IS95-A and ANSI J-STD-008 (referred to hereafter collectively as the IS-95 standard) promulgated by the telecommunication industry association (TIA) and used primarily within cellular telecommunications systems. The IS-95 standard incorporates code division multiple access (CDMA) signal modulation techniques to conduct multiple communications simultaneously over the same RF bandwidth. When combined with comprehensive power control, conducting multiple communications over the same bandwidth increases the total number of calls and other communications that can be conducted in a wireless communication system by, among other things, increasing the frequency reuse in comparison to other wireless telecommunication technologies. The use of CDMA techniques in a multiple access communication system is disclosed in U.S. Pat. No. 4,901,307, entitled “SPREAD SPECTRUM COMMUNICATION SYSTEM USING SATELLITE OR TERRESTRIAL REPEATERS,” and U.S. Pat. No. 5,103,459, entitled “SYSTEM AND METHOD FOR GENERATING SIGNAL WAVEFORMS IN A CDMA CELLULAR TELEPHONE SYSTEM,” both of which are assigned to the assignee of the present invention and incorporated by reference herein. 
     FIG. 1 provides a highly simplified illustration of a cellular telephone system configured in accordance with the use of the IS-95 standard. During operation, a set of subscriber units  10   a-d  conduct wireless communication by establishing one or more RF interfaces with one or more base stations  12   a-d  using CDMA-modulated RF signals. Each RF interface between a base station  12  and a subscriber unit  10  is comprised of a forward link signal transmitted from the base station  12 , and a reverse link signal transmitted from the subscriber unit. Using these RF interfaces, a communication with another user is generally conducted by way of mobile telephone switching office (MTSO)  14  and public switch telephone network (PSTN)  16 . The links between base stations  12 , MTSO  14  and PSTN  16  are usually formed via wire line connections, although the use of additional RF or microwave links is also known. 
     In accordance with the IS-95 standard, each subscriber unit  10  transmits user data via a single channel, non-coherent, reverse link signal at a maximum data rate of 9.6 or 14.4 kbits/sec depending on which rate set from a set of rate sets is selected. A non-coherent link is one in which phase information is not utilized by the received system. A coherent link is one in which the receiver exploits knowledge of the carrier signals phase during processing. The phase information typically takes the form of a pilot signal, but can also be estimated from the data transmitted. The IS-95 standard calls for a set of sixty-four Walsh codes, each comprised of sixty-four chips, to be used for the forward link. 
     The use of a single channel, non-coherent, reverse link signal having a maximum data rate of 9.6 of 14.4 kbits/sec as specified by IS-95 is well suited for a wireless cellular telephone system in which the typical communication involves the transmission of digitized voice or lower rate digital data, such as a facsimile. A non-coherent reverse link was selected because, in a system in which up to 80 subscriber units  10  may communicate with a base station  12  for each 1.2288 MHz of bandwidth allocated, providing the necessary pilot data in the transmission from each subscriber unit  10  would substantially increase the degree to which a set of subscriber units  10  interfere with one another. Also, at data rates of 9.6 or 14.4 kbits/sec, the ratio of the transmit power of any pilot data to the user data would be significant, and therefore also increase inter-subscriber unit interference. The use of a single channel reverse link signal was chosen because engaging in only one type of communication at a time is consistent with the use of wireline telephones, the paradigm on which current wireless cellular communications is based. Also, the complexity of processing a single channel is less than that associated with processing multiple channels. 
     As digital communications progress, the demand for wireless transmission of data for applications such as interactive file browsing and video teleconferencing is anticipated to increase substantially. This increase will transform the way in which wireless communications systems are used, and the conditions under which the associated RF interfaces are conducted. In particular, data will be transmitted at higher maximum rates and with a greater variety of possible rates. Also, more reliable transmission may become necessary as errors in the transmission of data are less tolerable than errors in the transmission of audio information. Additionally, the increased number of data types will create a need to transmit multiple types of data simultaneously. For example, it may be necessary to exchange a data file while maintaining an audio or video interface. Also, as the rate of transmission from a subscriber unit increases, the number of subscriber units  10  communicating with a base station  12  per amount of RF bandwidth will decrease, as the higher data transmission rates will cause the data processing capacity of the base station to be reached with fewer subscriber units  10 . In some instances, the current IS-95 reverse link may not be ideally suited for all these changes. Therefore, the present invention is related to providing a higher data rate, bandwidth efficient CDMA interface over which multiple types of communication can be performed. 
     SUMMARY 
     A novel and improved method and apparatus for high rate CDMA wireless communication is described. In accordance with one embodiment of the invention, a set of individually gain-adjusted subscriber channels are formed via the use of a set of orthogonal subchannel codes having a small number of PN spreading chips per orthogonal waveform period. Data to be transmitted via one of the transmit channels is low code rate error correction-encoded and sequence-repeated before being modulated with one of the subchannel codes, gain-adjusted, and summed with data modulated using the other subchannel codes. The resulting summed data is modulated using a user long code and a pseudorandom spreading code (PN code) and upconverted for transmission. The use of the short orthogonal codes provides interference suppression while still allowing extensive error correction coding and repetition for time diversity to overcome the Raleigh fading commonly experienced in terrestrial wireless systems. In the exemplary embodiment of the invention provided, the set of sub-channel codes are comprised of four Walsh codes, each orthogonal to the remaining set and four chips in duration. The use of a small number (e.g. four) sub-channels is preferred as it allows shorter orthogonal codes to be used. However, the use of a greater number of channels and therefore longer codes is consistent with the invention. In another embodiment of the invention the length, or number of chips, in each channel code is different to further reduced the peak-to-average transmit power. 
     In a preferred exemplary embodiment of the invention, pilot data is transmitted via a first one of the transmit channels and power control data transmitted via a second transmit channel. The remaining two transmit channels are used for transmitting non-specified digital data including user data or signaling data, or both. In an exemplary embodiment, one of the two non-specified transmit channels is configured for BPSK modulation and transmission over the quadrature channel. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The features, objects, and advantages of the present invention will become more apparent from the detailed description set forth below when taken in conjunction with the drawings in which like reference characters identify correspondingly throughout and wherein: 
     FIG. 1 is a block diagram of cellular telephone system; 
     FIG. 2 is a block diagram of a subscriber unit and base station configured in accordance with the exemplary embodiment of the invention; 
     FIG. 3 is a block diagram of a BPSK channel encoder and a QPSK channel encoder configured in accordance with the exemplary embodiment of the invention; 
     FIG. 4 is a block diagram of a transmit signal processing system configured in accordance with the exemplary embodiment of the invention; 
     FIG. 5 is a block diagram of a receive processing system configured in accordance with the exemplary embodiment of the invention; 
     FIG. 6 is a block diagram of a finger processing system configured in accordance with one embodiment of the invention; 
     FIG. 7 is a block diagram of a BPSK channel decoder and a QPSK channel decoder configured in accordance with the exemplary embodiment of the invention; and 
     FIG. 8 is a block diagram of a transmit signal processing system configured in accordance with a second exemplary embodiment of the invention; 
     FIG. 9 is a block diagram of a finger processing system configured in accordance with one embodiment of the invention; 
     FIG. 10 is a block diagram of a transmit signal processing system configured in accordance with another embodiment of the invention; 
     FIG. 11 is a block diagram of the coding performed for the fundamental channel when configured in accordance with one embodiment of the invention; 
     FIG. 12 is a block diagram of the coding performed for the fundamental channel when configured in accordance with one embodiment of the invention; 
     FIG. 13 is a block diagram of the coding performed for the supplemental channel when configured in accordance with one embodiment of the invention; and 
     FIG. 14 is a block diagram of the coding performed for the control channel when configured in accordance with one embodiment of the invention. 
    
    
     DETAILED DESCRIPTION 
     A novel and improved method and apparatus for high rate CDMA wireless communication is described in the context of the reverse link transmission portion of a cellular telecommunications system. While the invention is particularly adapted for use within the multipoint-to-point reverse link transmission of a cellular telephone system, the present invention is equally applicable to forward link transmissions. In addition, many other wireless communication systems will benefit by incorporation of the invention, including satellite based wireless communication systems, point-to-point wireless communication systems, and systems transmitting radio frequency signals via the use of co-axial or other broadband cables. 
     FIG. 2 is a block diagram of receive and transmit systems configured as a subscriber unit  100  and a base station  120  in accordance with one embodiment of the invention. A first set of data (BPSK data) is received by BPSK channel encoder  103 , which generates a code symbol stream configured for performing BPSK modulation that is received by modulator  104 . A second set of data (QPSK data) is received by QPSK channel encoder  102 , which generates a code symbol stream configured for performing QPSK modulation that is also received by modulator  104 . Modulator  104  also receives power control data and pilot data, which are modulated along with the BPSK and QPSK encoded data in accordance with code division multiple access (CDMA) techniques to generate a set of modulation symbols received by RF processing system  106 . RF processing system  106  filters and upconverts the set of modulation symbols to a carrier frequency for transmission to the base station  120  using antenna  108 . 
     While only one subscriber unit  100  is shown, multiple subscriber units communicate with base station  120  in the preferred embodiment. Within base station  120 , RF processing system  122  receives the transmitted RF signals by way of antenna  121  and performs bandpass filtering, downconversion to baseband, and digitization. Demodulator  124  receives the digitized signals and performs demodulation in accordance with CDMA techniques to produce power control, BPSK, and QPSK soft decision data. BPSK channel decoder  128  decodes the BPSK soft decision data received from demodulator  124  to yield a best estimate of the BPSK data, and QPSK channel decoder  126  decodes the QPSK soft decision data received by demodulator  124  to produce a best estimate of the QPSK data. The best estimate of first and second set of data is then available for further processing or forwarding to a next destination, and the received power control data used either directly, or after decoding, to adjust the transmit power of the forward link channel used to transmit data to subscriber unit  100 . 
     FIG. 3 is a block diagram of BPSK channel encoder  103  and QPSK channel encoder  102  when configured in accordance with the exemplary embodiment of the invention. Within BPSK channel encoder  103  the BPSK data is received by CRC check sum generator  130  which generates a check sum for each 20-ms frame of the first set of data. The frame of data along with the CRC check sum is received by tail bit generator  132  which appends tail bits comprised of eight logic zeros at the end of each frame to provide a known state at the end of the decoding process. The frame including the code tail bits and CRC check sum is then received by convolutional encoder  134  which performs, constraint length (K) 9, rate (R) ¼ convolutional encoding thereby generating code symbols at a rate four times the encoder input rate (E R ). In the alternative embodiment of the invention, other encoding rates are performed including rate ½, but the use of rate {fraction ( 1 / 4 )} is preferred due to its optimal complexity-performance characteristics. Block interleaver  136  performs bit interleaving on the code symbols to provide time diversity for more reliable transmission in fast fading environments. The resulting interleaved symbols are received by variable starting point repeater  138 , which repeats the interleaved symbol sequence a sufficient number of times N R  to provide a constant rate symbol stream, which corresponds to outputting frames having a constant number of symbols. Repeating the symbol sequence also increases the time diversity of the data to overcome fading. In the exemplary embodiment, the constant number of symbols is equal to 6,144 symbols for each frame making the symbol rate 307.2 kilosymbols per second (ksps). Also, variable starting point repeater  138  uses a different starting point to begin the repetition for each symbol sequence. When the value of NR necessary to generate 6,144 symbols per frame is not an integer, the final repetition is only performed for a portion of the symbol sequence. The resulting set of repeated symbols are received by BPSK mapper  139  which generates a BPSK code symbol stream (BPSK) of +1 and −1 values for performing BPSK modulation. In an alternative embodiment of the invention varible starting point repeater  138  is placed before block interleaver  136  so that block interleaver  136  receives the same number of symbols for each frame. 
     Within QPSK channel encoder  102  the QPSK data is received by CRC check sum generator  140  which generates a check sum for each 20-ms frame. The frame including the CRC check sum is received by code tail bits generator  142  which appends a set of eight tail bits of logic zeros at the end of the frame. The frame, now including the code tail bits and CRC check sum, is received by convolutional encoder  144  which performs K=9, R=¼ convolutional encoding, thereby generating symbols at a rate four times the encoder input rate (E R ). Block interleaver  146  performs bit interleaving on the symbols and the resulting interleaved symbols are received by variable starting point repeater  148 . Variable starting point repeater  148  repeats the interleaved symbol sequence a sufficient number of times N R  using a different starting point within the symbol sequence for each repetition to generate 12,288 symbols for each frame making the code symbol rate 614.4 kilosymbols per second (ksps). When N R  is not an integer, the final repetition is performed for only a portion of the symbol sequence. The resulting repeated symbols are received by QPSK mapper  149  which generates a QPSK code symbol stream configured for performing QPSK modulation comprised of an in-phase QPSK code symbol stream of +1 and −1 values (QPSK I ), and a quadrature-phase QPSK code symbol stream of +1 and −1 values (QPSK Q ). In an alternative embodiment of the invention variable staring point repeater  148  is placed before block interleaver  146  so that block interleaver  146  receives the same number of symbols for each frame. 
     FIG. 4 is a block diagram of modulator  104  of FIG. 2 configured in accordance with the exemplary embodiment of the invention. The BPSK symbols from BPSK channel encoder  103  are each modulated by Walsh code W 2  using a multiplier  150   b , and the QPSK I  and QPSK Q  symbols from QPSK channel encoder  102  are each modulated with Walsh code W 3  using multipliers  150   c  and  150   d . The power control data (PC) is modulated by Walsh code W 1  using multiplier  150   a . Gain adjust  152  receives pilot data (PILOT), which in the preferred embodiment of the invention is comprised of the logic level associated with positive voltage, and adjusts the amplitude according to a gain adjust factor A 0 . The PILOT signal provides no user data but rather provides phase and amplitude information to the base station so that it can coherently demodulate the data carried on the remaining sub-channels, and scale the soft-decision output values for combining. Gain adjust  154  adjusts the amplitude of the Walsh code W 1  modulated power control data according to gain adjust factor A 1 , and gain adjust  156  adjusts the amplitude of the Walsh code W 2  modulated BPSK channel data according amplification variable A 2 . Gain adjusts  158   a  and  b  adjust the amplitude of the in-phase and quadrature-phase Walsh code W 3  modulated QPSK symbols respectively according to gain adjust factor A 3 . The four Walsh codes used in the preferred embodiment of the invention are shown in Table I. 
     
       
         
               
               
               
             
           
               
                   
                 TABLE I 
               
               
                   
                   
               
               
                   
                   
                 Modulation 
               
               
                   
                 Walsh Code 
                 Symbols 
               
               
                   
                   
               
             
             
               
                   
                 W 0   
                 ++++ 
               
               
                   
                 W 1   
                 +−+− 
               
               
                   
                 W 2   
                 ++−− 
               
               
                   
                 W 3   
                 +−−+ 
               
               
                   
                   
               
             
          
         
       
     
     It will be apparent to one skilled in the art that the W 0  code is effectively no modulation at all, which is consistent with processing of the pilot data shown. The power control data is modulated with the W 1  code, the BPSK data with the W 2  code, and the QPSK data with the W 3  code. Once modulated with the appropriate Walsh code, the pilot, power control data, and BPSK data are transmitted in accordance with BPSK techniques, and the QPSK data (QPSK I  and QPSK Q ) in accordance with QPSK techniques as described below. It should also be understood that it is not necessary that every orthogonal channel be used, and that the use of only three of the four Walsh codes where only one user channel is provided is employed in an alternative embodiment of the invention. 
     The use of short orthogonal codes generates fewer chips per symbol, and therefore allows for more extensive coding and repetition when compared to systems incorporating the use of longer Walsh codes. This more extensive coding and repetition provides protection against Raleigh fading which is a major source of error in terrestrial communication systems. The use of other numbers of codes and code lengths is consistent with the present invention, however, the use of a larger set of longer Walsh codes reduces this enhanced protection against fading. The use of four-chip codes is considered optimal because four channels provide substantial flexibility for the transmission of various types of data as illustrated below while also maintaining short code length. 
     Summer  160  sums the resulting amplitude adjusted modulation symbols from gain adjusts  152 ,  154 ,  156  and  158   a  to generate summed modulation symbols  161 . PN spreading codes PNI and PNQ are spread via multiplication with long code  180  using multipliers  162   a  and  b.  The resulting pseudo-random code provided by multipliers  162   a  and  162   b  are used to modulate the summed modulation symbols  161 , and gain adjusted quadrature-phase symbols QPSK Q    163 , via complex multiplication using multipliers  164   a-d  and summers  166   a  and  b.  The resulting in-phase term X I  and quadrature-phase term X Q  are then filtered (filtering not shown), and upconverted to the carrier frequency within RF processing system  106  shown in a highly simplified form using multipliers  168  and an in-phase and a quadrature-phase sinusoid. An offset QPSK upconversion could also be used in an alternative embodiment of the invention. The resulting in-phase and quadrature-phase upconverted signals are summed using summer  170  and amplified by master amplifier  172  according to master gain adjust A M  to generate signal s(t) which is transmitted to base station  120 . In the preferred embodiment of the invention, the signal is spread and filtered to a 1.2288 MHz bandwidth to remain compatible with the bandwidth of existing CDMA channels. 
     By providing multiple orthogonal channels over which data may be transmitted, as well as by using variable rate repeaters that reduce the amount of repeating N R  performed in response to high input data rates, the above described method and system of transmit signal processing allows a single subscriber unit or other transmit system to transmit data at a variety of data rates. In particular, by decreasing the rate of repetition N R  performed by variable starting point repeaters  138  or  148  of FIG. 3, an increasingly higher encoder input rate E R  can be sustained. In an alternative embodiment of the invention rate ½ convolution encoding is performed with the rate of repetition N R  increased by two. A set of exemplary encoder rates E R  supported by various rates of repetition N R  and encoding rates R equal to ¼ and ½ for the BPSK channel and the QPSK channel are shown in Tables II and III respectively. 
     
       
         
               
             
               
               
               
               
               
               
             
               
               
               
               
               
               
             
           
               
                 TABLE II 
               
             
             
               
                   
               
               
                 BPSK Channel 
               
             
          
           
               
                   
                   
                 Encoder Out 
                 N R,R=1/4   
                 Encoder Out 
                 N R,R=1/2   
               
               
                   
                 E R,BPSK   
                 R = ¼ 
                 (Repetition 
                 R = ½ 
                 (Repetition 
               
               
                 Label 
                 (bps) 
                 (bits/frame) 
                 Rate, R = ¼) 
                 (bits/frame) 
                 Rate, R = ½) 
               
               
                   
               
             
          
           
               
                 High Rate-72 
                 76,800 
                 6,144 
                  1 
                 3,072 
                  2 
               
               
                 High Rate-64 
                 70,400 
                 5,632 
                  1{fraction (1/11)} 
                 2,816 
                  2{fraction (2/11)} 
               
               
                   
                 51,200 
                 4,096 
                  1½ 
                 2,048 
                  3 
               
               
                 High Rate-32 
                 38,400 
                 3,072 
                  2 
                 1,536 
                  4 
               
               
                   
                 25,600 
                 2,048 
                  3 
                 1,024 
                  6 
               
               
                 RS2-Full Rate 
                 14,400 
                 1,152 
                  5⅓ 
                 576 
                  10⅔ 
               
               
                 RS1-Full Rate 
                 9,600 
                 768 
                  8 
                 384 
                  16 
               
               
                 NULL 
                 850 
                 68 
                 90{fraction (6/17)} 
                 34 
                 180{fraction (12/17)} 
               
               
                   
               
             
          
         
       
     
     
       
         
               
             
               
               
               
               
               
               
             
               
               
               
               
               
               
             
           
               
                 TABLE III 
               
             
             
               
                   
               
               
                 QPSK Channel 
               
             
          
           
               
                   
                   
                 Encoder Out 
                 N R,R=1/4   
                 Encoder Out 
                 N R,R=1/2   
               
               
                   
                 E R,QPSK   
                 R = ¼ 
                 (Repetition 
                 R = ½ 
                 (Repetition 
               
               
                 Label 
                 (bps) 
                 (bits/frame) 
                 Rate, R = ¼) 
                 (bits/frame) 
                 Rate, R = ½) 
               
               
                   
               
             
          
           
               
                   
                 153,600 
                 12,288 
                  1 
                 6,144 
                  2 
               
               
                 High Rate-72 
                 76,800 
                 6,144 
                  2 
                 3,072 
                  4 
               
               
                 High Rate-64 
                 70,400 
                 5,632 
                  2 {fraction (2/11)} 
                 2,816 
                  4 {fraction (4/11)} 
               
               
                   
                 51,200 
                 4,096 
                  3 
                 2,048 
                  6 
               
               
                 High Rate-32 
                 38,400 
                 3,072 
                  4 
                 1,536 
                  8 
               
               
                   
                 25,600 
                 2,048 
                  6 
                 1,024 
                  12 
               
               
                 RS2-Full Rate 
                 14,400 
                 1,152 
                  10 ⅔ 
                 576 
                  21 ⅓ 
               
               
                 RS1-Full Rate 
                 9,600 
                 768 
                  16 
                 384 
                  32 
               
               
                 NULL 
                 850 
                 68 
                 180 {fraction (12/17)} 
                 34 
                 361 {fraction (7/17)} 
               
               
                   
               
             
          
         
       
     
     Tables II and III show that by adjusting the number of sequence repetitions N R , a wide variety of data rates can be supported including high data rates, as the encoder input rate E R  corresponds to the data transmission rate minus a constant necessary for the transmission of CRC, code tail bits and any other overhead information. As also shown by tables II and III, QPSK modulation may also be used to increase the data transmission rate. Rates expected to be used commonly are provided labels such as “High Rate-72” and “High Rate-32.” Those rates noted as High Rate-72, High Rate-64, and High Rate-32 have traffic rates of 72, 64 and 32 kbps, respectively, plus, these rates are multiplexed in signaling and other control data with rates of 3.6, 5.2, and 5.2 kbps, respectively, in the exemplary embodiment of the invention. Rates RS1-Full Rate and RS2-Full Rate correspond to rates used in IS-95 compliant communication systems, and therefore, are also expected to receive substantial use for purposes of compatibility. The null rate is the transmission of a single bit and is used to indicate a frame erasure, which is also part of the IS-95 standard. 
     The data transmission rate may also be increased by simultaneously transmitting data over two or more of the multiple orthogonal channels performed either in addition to, or instead of, increasing the transmission rate via reduction of the repetition rate N R . For example, a multiplexer (not shown) could split a single data source into a multiple data source to be transmitted over multiple data sub-channels. Thus, the total transmit rate can be increased via either transmission over a particular channel at higher rates, or multiple transmissions performed simultaneously over multiple channels, or both, until the signal processing capability of the receive system is exceeded and the error rate becomes unacceptable, or the maximum transmit power of the transmit system power is reached. 
     Providing multiple channels also enhances flexibility in the transmission of different types of data. For example, the BPSK channel may be designated for voice information and the QPSK channel designated for transmission of digital data. This embodiment could be more generalized by designating one channel for transmission of time-sensitive data such as voice at a lower data rate, and designating the other channel for transmission of less time sensitive data such as digital files. In this embodiment interleaving could be performed in larger blocks for the less time sensitive data to further increase time diversity. In another embodiment of the invention, the BPSK channel performs the primary transmission of data, and the QPSK channel performs overflow transmission. The use of orthogonal Walsh codes eliminates or substantially reduces any interference among the set of channels transmitted from a subscriber unit, and thus minimizes the transmit energy necessary for their successful reception at the base station. 
     To increase the processing capability at the receive system, and therefore increase the extent to which the higher transmission capability of the subscriber unit may be utilized, pilot data is also transmitted via one of the orthogonal channels. Using the pilot data, coherent processing can be performed at the receive system by determining and removing the phase offset of the reverse link signal. Also, the pilot data can be used to optimally weigh multipath signals received with different time delays before being combined in a rake receiver. Once the phase offset is removed, and the multipath signals properly weighted, the multipath signals can be combined decreasing the power at which the reverse link signal must be received for proper processing. This decrease in the required receive power allows greater transmissions rates to be processed successfully, or conversely, the interference between a set of reverse link signals to be decreased. While some additional transmit power is necessary for the transmission of the pilot signal, in the context of higher transmission rates, the ratio of pilot channel power to the total reverse link signal power is substantially lower than that associated with lower data rate digital voice data transmission cellular systems. Thus, within a high data rate CDMA system the E b /N 0  gains achieved by the use of a coherent reverse link outweigh the additional power necessary to transmit pilot data from each subscriber unit. 
     The use of gain adjusts  152 - 158 , as well as master amplifier  172 , further increases the degree to which the high transmission capability of the above described system can be utilized by allowing the transmit system to adapt to various radio channel conditions, transmission rates, and data types. In particular, the transmit power of a channel that is necessary for proper reception may change over time, and with changing conditions, in a manner that is independent of the other orthogonal channels. For example, during the initial acquisition of the reverse link signal the power of the pilot channel may need to be increased to facilitate detection and synchronization at the base station. Once the reverse link signal is acquired, however, the necessary transmit power of the pilot channel would substantially decrease and would vary depending on various factors including the subscriber units rate of movement. Accordingly, the value of the gain adjust factor AO would be increased during signal acquisition, and then reduced during an ongoing communication. In another example, when information more tolerable of error is being transmitted via the forward link, or the environment in which the forward link transmission is taking place is not prone to fade conditions, the gain adjust factor A 1  may be reduced as the need to transmit power control data with a low error rate decreases. In one embodiment of the invention, whenever power control adjustment is not necessary the gain adjust factor A 1  is reduced to zero. 
     In another embodiment of the invention, the ability to gain adjust each orthogonal channel or the entire reverse link signal is further exploited by allowing the base station  120  or other receive systems to alter the gain adjust of a channel, or of the entire reverse link signal, via the use of power control commands transmitted via the forward link signal. In particular, the base station may transmit power control information requesting the transmit power of a particular channel or the entire reverse link signal be adjusted. This is advantageous in many instances including when two types of data having different sensitivity to error, such as digitized voice and digital data, are being transmitted via the BPSK and QPSK channels. In this case, the base station  120  would establish different target error rates for the two associated channels. If the actual error rate of a channel exceeded the target error rate, the base station would instruct the subscriber unit to reduce the gain adjust of that channel until the actual error rate reached the target error rate. This would eventually lead to the gain adjust factor of one channel being increased relative to the other. That is, the gain adjust factor associated with the more error sensitive data would be increased relative to the gain adjust factor associated with the less sensitive data. In other instances, the transmit power of the entire reverse link may require adjustment due to fade conditions or movement of the subscriber unit  100 . In these instances, the base station  120  can do so via transmission of a single power control command. 
     Thus, by allowing the gain of the four orthogonal channels to be adjusted independently, as well as in conjunction with one another, the total transmit power of the reverse link signal can be kept at the minimum necessary for successful transmission of each data type, whether it is pilot data, power control data, signaling data, or different types of user data. Furthermore, successful transmission can be defined differently for each data type. Transmitting with the minimum amount of power necessary allows the greatest amount of data to be transmitted to the base station given the finite transmit power capability of a subscriber unit, and also reduces the interfere between subscriber units. This reduction in interference increases the total communication capacity of the entire CDMA wireless cellular system. 
     The power control channel used in the reverse link signal allows the subscriber unit to transmit power control information to the base station at a variety of rates including a rate of 800 power control bits per second. In the preferred embodiment of the invention, a power control bit instructs the base station to increase or decrease the transmit power of the forward link traffic channel being used to transmit information to the subscriber unit. While it is generally useful to have rapid power control within a CDMA system, it is especially useful in the context of higher data rate communications involving data transmission, because digital data is more sensitive to errors, and the high transmission causes substantial amounts of data to be lost during even brief fade conditions. Given that a high speed reverse link transmission is likely to be accompanied by a high speed forward link transmission, providing for the rapid transmission of power control over the reverse link further facilitates high speed communications within CDMA wireless telecommunications systems. 
     In an alternative exemplary embodiment of the invention a set of encoder input rates E R  defined by the particular N R  are used to transmit a particular type of data. That is, data may be transmitted at a maximum encoder input rate E R  or at a set of lower encoder input rates E R , with the associated N R  adjusted accordingly. In the preferred implementation of this embodiment, the maximum rates corresponds to the maximum rates used in IS-95 compliant wireless communication system, referred to above with respect to Tables II and III as RS1-Full Rate and RS2-Full Rate, and each lower rate is approximately one half the next higher rate, creating a set of rates comprised of a full rate, a half rate, a quarter rate, and an eighth rate. The lower data rates are preferable generated by increasing the symbol repetition rate N R  with value of N R  for rate set one and rate set two in a BPSK channel provided in Table IV. 
     
       
         
               
             
               
               
               
               
               
               
             
               
               
               
               
               
               
             
           
               
                 TABLE IV 
               
             
             
               
                   
               
               
                 RS1 and RS2 Rate Sets in BPSK Channel 
               
             
          
           
               
                   
                   
                 Encoder Out 
                 N R,R=1/4   
                 Encoder Out 
                 N R,R=1/2   
               
               
                   
                 E R,QPSK   
                 R = ¼ 
                 (Repetition 
                 R = ½ 
                 (Repetition 
               
               
                 Label 
                 (bps) 
                 (bits/frame) 
                 Rate, R = ¼) 
                 (bits/frame) 
                 Rate, R = ½) 
               
               
                   
               
             
          
           
               
                 RS2-Full Rate 
                 14,400 
                 1,152 
                  5 ⅓ 
                 576 
                  10⅔ 
               
               
                 RS2-Half Rate 
                 7,200 
                 576 
                 10⅔ 
                 288 
                  21⅓ 
               
               
                 RS2-Quarter 
                 3,600 
                 288 
                 21⅓ 
                 144 
                  42⅔ 
               
               
                 Rate 
               
               
                 RS2-Eighth 
                 1,900 
                 152 
                 40{fraction (8/19)} 
                 76 
                  80{fraction (16/19)} 
               
               
                 Rate 
               
               
                 RS1-Full Rate 
                 9,600 
                 768 
                  8 
                 384 
                  16 
               
               
                 RS1-Half Rate 
                 4,800 
                 384 
                 16 
                 192 
                  32 
               
               
                 RS1-Quarter 
                 2,800 
                 224 
                 27{fraction (3/7)} 
                 112 
                  54{fraction (6/7)} 
               
               
                 Rate 
               
               
                 RS1-Eighth 
                 1,600 
                 128 
                 48 
                 64 
                  96 
               
               
                 Rate 
               
               
                 NULL 
                 850 
                 68 
                 90{fraction (6/17)} 
                 34 
                 180{fraction (12/17)} 
               
               
                   
               
             
          
         
       
     
     The repetition rates for a QPSK channel is twice that for the BPSK channel. 
     In accordance with the exemplary embodiment of the invention, when the data rate of a frame changes with respect to the previous frame the transmit power of the frame is adjusted according to the change in transmission rate. That is, when a lower rate frame is transmitted after a higher rate frame, the transmit power of the transmit channel over which the frame is being transmitted is reduced for the lower rate frame in proportion to the reduction in rate, and vice versa. For example, if the transmit power of a channel during the transmission of a full rate frame is transmit power T, the transmit power during the subsequent transmission of a half rate frame is transmit power T/2. The reduction in transmit power is preferably performed by reducing the transmit power for the entire duration of the frame, but may also be performed by reducing the transmit duty cycle such that some redundant information is “blanked out.” In either case, the transmit power adjustment takes place in combination with a closed loop power control mechanism whereby the transmit power is further adjusted in response to power control data transmitted from the base station. 
     FIG. 5 is a block diagram of RF processing system  122  and demodulator  124  of FIG. 2 configured in accordance with the exemplary embodiment of the invention. Multipliers  180   a  and  180   b  downconvert the signals received from antenna  121  with an in-phase sinusoid and a quadrature phase sinusoid producing in-phase receive samples R I  and quadrature-phase receive samples R Q  receptively. It should be understood that RF processing system  122  is shown in a highly simplified form, and that the signals are also match-filtered and digitized (not shown) in accordance with widely known techniques. Receive samples R I  and R Q  are then applied to finger demodulators  182  within demodulator  124 . Each finger demodulator  182  processes an instance of the reverse link signal transmitted by subscriber unit  100 , if such an instance is available, where each instance of the reverse link signal is generated via multipath phenomenon. While three finger demodulators are shown, the use of alternative numbers of finger processors are consistent with the invention including the use of a single finger demodulator  182 . Each finger demodulator  182  produces a set of soft decision data comprised of power control data, BPSK data, and QPSK I  data and QPSK Q  data. Each set of soft decision data is also time-adjusted within the corresponding finger demodulator  182 , although time adjustment could be performed within combiner  184  in an alternative embodiment of the invention. Combiner  184  then sums the sets of soft decision data received from finger demodulators  182  to yield a single instance of power control, BPSK, QPSK I  and QPSK Q  soft decision data. 
     FIG. 6 is a block diagram of a finger demodulator  182  of FIG. 5 configured in accordance with the exemplary embodiment of the invention. The RI and RQ receive samples are first time adjusted using time adjust  190  in accordance with the amount of delay introduced by the transmission path of the particular instance of the reverse link signal being processed. Long code  200  is mixed with pseudorandom spreading codes PNI and PNQ using multipliers  201 , and the complex conjugate of the resulting long code modulated PNI and PNQ spreading codes are complex-multiplied with the time adjusted RI and RQ receive samples using multipliers  202  and summers  204  yielding terms XI and XQ. Three separate instances of the XI and XQ terms are then demodulated using the Walsh codes W1, W2 and W3 respectively, and the resulting Walsh demodulated data is summed over four demodulation chips using 4 to 1 summers  212 . A fourth instance of the XI and XQ data is summed over four demodulation chips using summers  208 , and then filtered using pilot filters  214 . In the preferred embodiment of the invention pilot filter  214  performs averaging over a series of summations performed by summers  208 , but other filtering techniques will be apparent to one skilled in the art. The filtered in-phase and quadrature-phase pilot signals are used to phase rotate and scale the W1 and W2 Walsh code demodulated data in accordance with BPSK modulated data via complex conjugate multiplication using multipliers  216  and adders  217  yielding soft decision power control and BPSK data. The W3 Walsh code-modulated data is phase rotated using the in-phase and quadrature-phase filtered pilot signals in accordance with QPSK modulated data using multipliers  218  and adders  220 , yielding soft decision QPSK data. The soft decision power control data is summed over 384 modulation symbols by 384 to 1 summer  222  yielding power control soft decision data. The phase rotated W2 Walsh code modulated data, the W3 Walsh code modulated data, and the power control soft decision data are then made available for combining. In an alternative embodiment of the invention, encoding and decoding is performed on the power control data as well. 
     In addition to providing phase information the pilot may also be used within the receive system to facilitate time tracking. Time tracking is performed by also processing the received data at one sample time before (early), and one sample time after (late), the present receive sample being processed. To determine the time that most closely matches the actual arrival time, the amplitude of the pilot channel at the early and late sample time can be compared with the amplitude at the present sample time to determine that which is greatest. If the signal at one of the adjacent sample times is greater than that at the present sample time, the timing can be adjusted so that the best demodulation results are obtained. 
     FIG. 7 is a block diagram of BPSK channel decoder  128  and QPSK channel decoder  126  (FIG. 2) configured in accordance with the exemplary embodiment of the invention. BPSK soft decision data from combiner  184  (FIG. 5) is received by accumulator  240  which stores the first sequence of 6,144/N R  demodulation symbols in the received frame where N R  depends on the transmission rate of the BPSK soft decision data as described above, and adds each subsequent set of 6,144/N R  demodulated symbols contained in the frame with the corresponding stored accumulated symbols. Block deinterleaver  242  deinterleaves the accumulated soft decision data from variable starting point accumlator  240 , and Viterbi decoder  244  decodes the deinterleaved soft decision data to produce hard decision data as well as CRC check sum results. Within QPSK decoder  126  QPSK I  and QPSK Q  soft decision data from combiner  184  (FIG. 5) are demultiplexed into a single soft decision data stream by demux  246  and the single soft decision data stream is received by accumulator  248  which accumulates every 6,144/N R  demodulation symbols where N R  depends on the transmission rate of the QPSK data. Block deinterleaver  250  deinterleaves the soft decision data from variable starting point accumulator  248 , and Viterbi decoder  252  decodes the deinterleaved modulation symbols to produce hard decision data as well as CRC check sum results. In the alternative exemplary embodiment described above with respect to FIG. 3 in which symbol repetition is performed before interleaving, accumulators  240  and  248  are placed after block deinterleavers  242  and  250 . In the embodiment of the invention incorporating the use of rate sets, and therefore in which the rate of particular frame is not known, multiple decoders are employed, each operating at a different transmission rate, and then the frame associated with the transmission rate most likely to have been used is selected based on the CRC checksum results. The use of other error checking methods is consistent with the practice of the present invention. 
     FIG. 8 is a block diagram of modulator  104  (FIG. 2) configured in an alternative embodiment of the invention in which a single BPSK data channel is employed. Pilot data is gain adjusted by gain adjust  452  in accordance with gain adjust factor A 0 . Power control data is modulated with Walsh code W 1  by multiplier  150   a  and gain adjusted by gain adjust  454  in accordance with gain adjust factor A 1 . The gain adjusted pilot data and power control data are summed by summer  460  producing summed data  461 . BPSK data is modulated with Walsh code W 2  by multiplier  150   b  and then gain adjusted using gain adjust  456  in accordance with gain adjust factor A 2 . 
     In-phase pseudo random spreading code (PN I ) and quadrature-phase pseudo random spreading code (PN Q ) are both modulated with long code  480 . The resulting long code modulated PN I  and PN Q  codes are complex multiplied with the summed data  461  and the gain adjusted BPSK data from gain adjust  456  using multipliers  464   a-d  and summers  466   a-b  yielding terms X I  and X Q . Terms X I  and X Q  are then upconverted with in-phase and quadrature-phase sinusoids suing multipliers  468  and the resulting upconverted signals are summed by summers  470  respectively, and amplified by amplifier  472  in accordance with amplitude factor AM generating signal s(t). 
     The embodiment shown in FIG. 8 differs from the other embodiments described herein in that the BPSK data is placed in the quadrature-phase channel while the pilot data and power control data are placed in the in-phase channel. In the previous embodiments of the invention described herein the BPSK data is placed the in-phase channel along with the pilot data and power control data. Placing the BPSK data in the quadrature-phase channel and the pilot and power control data in the in-phase channel reduces the peak-to-average power ratio of the reverse link signal the phases of the channels are orthogonal causing the magnitude of the sum of the two channels to vary less in response to changing data. This reduces the peak power required to maintain a given average power, and thus reduces the peak-to-average power ratio characteristic of the reverse link signal. This reduction in the peak-to-average power ratio decreases the peek power at which a reverse link signal must be received at the base station in order to sustain a given transmission rate, and therefore increases the distance in which a subscriber unit having a maximum transmit power may be located from the base station before it is unable to transmit a signal that can received at base station with the necessary peek power. This increases the range at which the subscriber unit can successfully conduct communication at any given data rate, or alternatively allows greater data rates to be sustained at a given distance. 
     FIG. 9 is a block diagram of finger demodulator  182  when configured in accordance with the embodiment of the invention shown in FIG.  8 . Receive samples R I  and R Q  are time adjusted by timing adjust  290  and the PN I  and PN Q  codes are multiplied by long code  200  using multipliers  301 . The time adjusted receive samples are then multiplied by the complex conjugate of the PN I  and PN Q  codes using multipliers  302  and summers  304  yielding terms X I  and X Q . A first and second instance of the X I  and X Q  terms are demodulated using Walsh code W 1  and Walsh code W 2  using multipliers  310  and the resulting demodulation symbols are summed in sets of four using summers  312 . A third instance of the X I  and X Q  terms are summed over four demodulation symbols by summers  308  to generate pilot reference data. The pilot reference data is filtered by pilot filters  314  and used to phase rotate and scale the summed Walsh code modulated data using multipliers  316  and adders  320  producing BPSK soft decision data, and after being summed over 384 symbols by 384:1 summer  322 , soft decision power control data. 
     FIG. 10 is a block diagram of a transmit system configured in accordance with still another embodiment of the invention. Channel gain  400  gain adjusts pilot channel  402  based on gain variable A 0 . Fundamental channel symbols  404  are mapped into +1 and −1 values by mapper  405 , and each symbol is modulated with Walsh code W F  equal to +,+,−,− (where +=+1 and −=−1). The W F  modulated data is gain-adjusted based on gain variable A 1  by gain channel adjust  406 . The outputs of gain adjusts  400  and  406  are summed by summer  408  yielding in-phase data  410 . 
     Supplemental channel symbols  411  are mapped to + and − values by signal point mapper  412 , and each symbol is modulated with a Walsh code W S  equal to +,−. Channel gain adjust  414  adjusts the gain of the W S  modulated data. Control channel data  415  is mapped to + and − values by mapper  416 . Each symbol is modulated with a Walsh code W C  equal to +, +, +, +, −, −, −, −. The W C  modulated symbols are gain-adjusted by channel gain adjust  418  based on gain variable A 3 , and the output of channel gain adjusts  414  and  418  are summed by summer  419  to produce quadrature phase data  420 . 
     It should be apparent that, since the Walsh codes W F  and W S  are different lengths, and are generated at the same chip rate, the fundamental channel transmits data symbols at a rate that is half that of the supplemental channel. For similar reasons, it should be apparent that the control channel transmits data symbols at half the rate of the fundamental channel. 
     In-phase data  410  and quadrature phase data  420  are complex multiplied by the PN I  and PN Q  spreading codes as shown, yielding in-phase term X I  and quadrature phase term X Q . The quadrature phase term X Q  is delay by ½ the duration of a PN spreading code chip to perform offset QPSK spreading, and then term X I  and term X Q  are upconverted in accordance with the RF processing system  106  shown in FIG. 4, and described above. 
     By using Walsh codes W F , W S  and W C  having different lengths as described above, this alternative embodiment of the invention provides a set of communication channels having a greater variety of rates. Additionally, the use of a shorter, two-chip, Walsh code W S  for the supplemental channel provides an orthogonal higher data rate supplemental channel with a peak-to-average transmit power ratio that is less than that associated with the use of two channels based on 4-chip Walsh codes. This further enhances the performance of the transmit system in that a given amplifier will be able to sustain a higher rate, or transmit with greater range, using the lower peak-to-average transmit power waveform. 
     The Walsh code allocation scheme described with regard to FIG. 10, can also be viewed as the allocation of eight-chip Walsh space in accordance with Table VI. 
     
       
         
               
               
               
               
             
               
               
               
               
             
           
               
                   
                 TABLE VI 
               
               
                   
                   
               
               
                   
                 Eight-Chip Walsh 
                   
                   
               
               
                   
                 Code 
                   
                 Channel 
               
               
                   
                   
               
             
             
               
                   
               
             
          
           
               
                   
                 ++++ 
                 ++++ 
                 Pilot 
               
               
                   
                 +−+− 
                 +−+− 
                 Supplemental 
               
               
                   
                 ++−− 
                 ++−− 
                 Fundamental 
               
               
                   
                 +−−+ 
                 +−−+ 
                 Supplemental 
               
               
                   
                 ++++ 
                 −−−− 
                 Control 
               
               
                   
                 +−+− 
                 −+−+ 
                 Supplemental 
               
               
                   
                 ++−− 
                 −−++ 
                 Fundamental 
               
               
                   
                 +−−+ 
                 −++− 
                 Supplemental 
               
               
                   
                   
               
             
          
         
       
     
     In addition to reducing the peak to average transmit power ratio, allocating sets of eight-chip Walsh channels using a single shorter Walsh code decreases the complexity of the transmit system. For example, modulating with four eight-chip Walsh codes and summing the results require additional circuitry and therefore would be more complex. 
     It is further contemplated that the transmission system shown in FIG. 10 can operate at various spreading bandwidths, and therefore with the Walsh codes and spreading codes generated at various rates other than 1.2288 Mchips/second. In particular, a spreading bandwidth of 3.6864 MHz is contemplated, with a corresponding Walsh and spreading code rate of 3.6864 Mchips/second. FIGS. 11-14 illustrate the coding performed for the fundamental, supplemental and control channels in accordance with the use of a 3.6864-MHz spreading bandwidth. Typically, to adjust the coding for use with a 1.2288-MHz spreading bandwidth the number of symbol repeats is reduced. This principal or adjusting the number of symbol repeats can be applied more generally to increases in the spreading bandwidth including, for example, the use of a 5-MHz spreading bandwidth. Adjustments performed to the coding for a 1.2288-MHz spreading bandwidth system other than reduction in the number of symbol repeats are particularly noted in the description of FIGS. 11-14 provided below. 
     FIG. 11 shows the coding performed for the four rates (i.e. full, half, quarter and eight rate) that make up the IS-95 rate set 1 when performed in accordance with one embodiment of the invention. Data is supplied in 20-ms frames having the number of bits shown for each rate, and CRC check bits and eight tail bits are added by CRC checks sum generators  500   a-d  and tail bit generators  502   a-d.  Additionally, rate ¼ convolutional encoding is performed for each rate by convolutional encoders  504   a-d,  generating four code symbols for each data bit, CRC bit, or tail bit. The resulting frame of code symbols is block interleaved using block interleavers  506   a-d,  generating the number of symbols indicated. For the lower three rates, the symbols are transmitted repeatedly by transmission repeaters  508   a-c , as indicated, causing 768 code symbols to be generated for each frame. The 768 code symbols for each rate are then repeated 24 times by symbol repeaters  510   a-d  generating 18,432 code symbols per frame for each rate. 
     As discussed above, each code symbol in the fundamental channel is modulated with a four bit Walsh code W F  generated at 3,686,400 chips per second (3.6864 Mchips/second). Thus, for a 20-ms time interval ({fraction (1/50)} th  of a second) the number of Walsh and spreading code chips is 73,728, which corresponds to 4 Walsh chips for each of the 18,432 code symbol in the frame. 
     For a system operating at 1.2288 Mchips/second, the number of symbol repeats performed by symbol repeaters  510   a-d  is reduced to eight (8). Additionally, transmission repeater  508   b  repeats the sequence of symbols in the frame three (3) times, plus 120 of the symbols are transmitted a fourth time, and transmission repeater  508   c  repeats the sequence of symbols in the frame six (6) times, plus 48 of the symbols are repeated a seventh time. Additionally, a fourth transmission repeater (or fourth transmission repeat step) is included for the full rate (not shown) which transmits 384 of the sequence of symbols contained in the frame a second time. These repeated transmissions all provide 768 symbols of data which, when repeated eight times by symbol repeaters  510   a-d,  correspond to 6,144 symbols, which is the number of chips in a 20 ms frame at 1.2288 Mchips/second. 
     FIG. 12 shows the coding performed for the four rates that make up IS-95 rate set 2 when performed in accordance with one embodiment of the invention. Data is supplied in 20 ms frames having the number of bits shown for each rate, and a reserve bit is added by reserve bit augmenters  521   a-d  for each rate. CRC check bits and eight tail bits are also added by CRC checks sum generators  520   a-d  and tail bit generators  522   a-d.  Additionally, rate ¼ convolutional encoding is performed for each rate by convolutional encoders  524   a-d,  generating four code symbols for each data, CRC or tail bit. The resulting frame of code symbols is block interleaved using block interleaves  526   a-d  generating the number of symbols indicated. For the lower three rates, the symbols are transmitted repeatedly by transmission repeaters  528   a-c  as indicated, causing 768 code symbols to be generated for each frame. The 768 code symbols for each rate are then repeated 24 times by symbol repeaters  530   a-d  generating 18,432 code symbols per frame for each rate. 
     For a system operating at 1.2288 MHz spreading bandwidth, the number of symbol repeats performed by symbol repeaters  530   a-d  is reduced to four (4). Additionally, transmission repeater  528   a  transmits the sequence of symbols in the frame two (2) times, plus 384 of the symbols are transmitted a third time. Transmission repeater  528   b  repeats the sequence of symbols in the frame five (5) times, plus 96 of the symbols are transmitted a sixth time. Transmission repeater  528   c  repeats the sequence of symbols in the frame ten (10) times, plus 96 of the symbols are repeated an eleventh time. Additionally, a fourth transmission repeater (or fourth transmission repeat step) is included for the full rate (not shown) which transmits  384  of the sequence of symbols contained in the frame a second time. These repeated transmissions all provide 1,536 symbols of data which, when repeated four times by symbol repeaters  530   a-d,  correspond to 6,144 symbols. 
     FIG. 13 illustrates the coding performed for the supplemental channel when performed in accordance with one embodiment of the invention. Frames of data are supplied at any of the eleven rates indicated, and CRC check sum generator  540  adds 16 bits of CRC checksum data. Tail bit generator  542  adds eight bits of encoder tail data resulting in frames having the data rates shown. Convolution encoder  544  performs rate ¼, constraint length K=9, encoding generating for code symbols for each data, CRC or tail bit received, and block interleaver  546  performs block interleaving on each frame, and outputs the number of code symbols shown for each frame in accordance with the input frame size. Symbol repeater  548  repeats the frames N times depending on the input frame size as indicated. 
     The encoding for an additional twelfth rate is shown, which is performed in a similar fashion to the eleven rates, with the exception that rate ½ encoding is performed instead of rate ¼. Additionally, no symbol repetition is performed. 
     A list of frame sizes, encoder input rates, code rates and symbol repetition factors N for various chip rates that can be applied to FIG. 13 to adjust for different chip rates (which correspond to spreading bandwidths) is provided in Table VII. 
     
       
         
               
               
               
               
               
             
               
               
               
               
               
             
           
               
                 TABLE VII 
               
               
                   
               
               
                   
                   
                 Encoder 
                   
                 Symbol 
               
               
                 Chip 
                 Number 
                 Input 
                   
                 Repetition 
               
               
                 Rate 
                 of Octets 
                 Rate 
                 Code 
                 Factor 
               
               
                 (Mcps) 
                 per Frame 
                 (kbps) 
                 Rate 
                 (N) 
               
               
                   
               
             
             
               
                   
               
             
          
           
               
                 1.2288 
                 21 
                 9.6 
                 1/4 
                 16 
               
               
                 1.2288 
                 45 
                 19.2 
                 1/4 
                 8 
               
               
                 1.2288 
                 93 
                 38.4 
                 1/4 
                 4 
               
               
                 1.2288 
                 189 
                 76.8 
                 1/4 
                 2 
               
               
                 1.2288 
                 381 
                 153.6 
                 1/4 
                 1 
               
               
                 1.2288 
                 765 
                 307.2 
                 1/2 
                 1 
               
               
                 3.6864 
                 21 
                 9.6 
                 1/4 
                 48 
               
               
                 3.6864 
                 33 
                 14.4 
                 1/4 
                 32 
               
               
                 3.6864 
                 45 
                 19.2 
                 1/4 
                 24 
               
               
                 3.6864 
                 69 
                 28.8 
                 1/4 
                 16 
               
               
                 3.6864 
                 93 
                 38.4 
                 1/4 
                 12 
               
               
                 3.6864 
                 141 
                 57.6 
                 1/4 
                 8 
               
               
                 3.6864 
                 189 
                 76.8 
                 1/4 
                 6 
               
               
                 3.6864 
                 285 
                 115.2 
                 1/4 
                 4 
               
               
                 3.6864 
                 381 
                 153.6 
                 1/4 
                 3 
               
               
                 3.6864 
                 573 
                 230.4 
                 1/4 
                 2 
               
               
                 3.6864 
                 1,149 
                 460.8 
                 1/4 
                 1 
               
               
                 3.6864 
                 2,301 
                 921.6 
                 1/2 
                 1 
               
               
                 7.3728 
                 21 
                 9.6 
                 1/4 
                 96 
               
               
                 7.3728 
                 33 
                 14.4 
                 1/4 
                 64 
               
               
                 7.3728 
                 45 
                 19.2 
                 1/4 
                 48 
               
               
                 7.3728 
                 69 
                 28.8 
                 1/4 
                 32 
               
               
                 7.3728 
                 93 
                 38.4 
                 1/4 
                 24 
               
               
                 7.3728 
                 141 
                 57.6 
                 1/4 
                 16 
               
               
                 7.3728 
                 189 
                 76.8 
                 1/4 
                 12 
               
               
                 7.3728 
                 285 
                 115.2 
                 1/4 
                 8 
               
               
                 7.3728 
                 381 
                 153.6 
                 1/4 
                 6 
               
               
                 7.3728 
                 573 
                 230.4 
                 1/4 
                 4 
               
               
                 7.3728 
                 765 
                 307.2 
                 1/4 
                 3 
               
               
                 7.3728 
                 1,149 
                 460.8 
                 1/4 
                 2 
               
               
                 7.3728 
                 2,301 
                 921.6 
                 1/4 
                 1 
               
               
                 7.3728 
                 4,605 
                 1,843.2 
                 1/2 
                 1 
               
               
                 14.7456 
                 21 
                 9.6 
                 1/4 
                 192 
               
               
                 14.7456 
                 33 
                 14.4 
                 1/4 
                 128 
               
               
                 14.7456 
                 45 
                 19.2 
                 1/4 
                 96 
               
               
                 14.7456 
                 69 
                 28.8 
                 1/4 
                 64 
               
               
                 14.7456 
                 93 
                 38.4 
                 1/4 
                 48 
               
               
                 14.7456 
                 141 
                 57.6 
                 1/4 
                 32 
               
               
                 14.7456 
                 189 
                 76.8 
                 1/4 
                 24 
               
               
                 14.7456 
                 285 
                 115.2 
                 1/4 
                 16 
               
               
                 14.7456 
                 381 
                 153.6 
                 1/4 
                 12 
               
               
                 14.7456 
                 573 
                 230.4 
                 1/4 
                 8 
               
               
                 14.7456 
                 765 
                 307.2 
                 1/4 
                 6 
               
               
                 14.7456 
                 1,149 
                 460.8 
                 1/4 
                 4 
               
               
                 14.7456 
                 1,533 
                 614.4 
                 1/4 
                 3 
               
               
                 14.7456 
                 2,301 
                 921.6 
                 1/4 
                 2 
               
               
                 14.7456 
                 4,605 
                 1,843.2 
                 1/4 
                 1 
               
               
                 14.7456 
                 9,213 
                 3,686.4 
                 1/2 
                 1 
               
               
                   
               
             
          
         
       
     
     FIG. 14 is a block diagram of the processing performed for the control channel for a 3.6864 MHz spreading bandwidth system. The processing is substantially similar to that associated with the other channels, except for the addition of a mux  560  and symbol repeater  562 , which operate to introduce uncoded power control bits into the code symbol stream. The power control bits are generated at a rate of 16 per frame, and repeated 18 times by symbol repeater  562  resulting in 288 power control bits per frame. The 288 power control bits are multiplexed into the frame of code symbols at a ratio of three power control bits per coded data symbol, generating 384 total symbols per frame. Symbol repeater  564  repeats the 384 bits 24 times generating 9,216 symbols per frame for an effective data rate of 500 kbits/second for the control data, and 800 kbits/second for the power control bits. The preferred processing performed for a 1.2288 MHz bandwidth system simply reduces the number of symbol repetitions performed from 24 to 8. 
     Thus, a multi-channel, high rate, CDMA wireless communication system has been described. The description is provided to enable any person skilled in the art to make or use the present invention. The various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without the use of the inventive faculty. Thus, the present invention is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.