Abstract:
An Orthogonal Frequency Division Multiplexing (OFDM) receiver that employs N second-order phase-lock loops sharing a common integrator (where N is the number of pilots in the system). The N second order phase-lock loops track out independent pilot phase rotations to facilitate the constructive averaging of the pilots&#39; phase information. At the same time, by sharing a common integrator, the OFDM receiver takes advantage of noise averaging over multiple pilots to obtain a cleaner frequency offset estimation. The OFDM receiver may also compensate for FFT window drift by calculating a phase difference between a selected pair of pilots and tracking the rate of change of the calculated phase difference over time. The calculated phase difference is used to control the position of an upstream FFT window after a predetermined phase difference threshold is exceeded. The tracked rate of change is used to continuously adjust the phase of downstream equalizer taps.

Description:
This application claims the benefit, under 35 U.S.C. §365 of International Application PCT/US01/20054, filed Jun. 22, 2001, which was published in accordance with PCT Article 21(2) on Jan. 3, 2003 in English. 
     BACKGROUND OF THE INVENTION 
     The present invention relates to processing orthogonal frequency division multiplexed (OFDM) signals. 
     A wireless LAN (WLAN) is a flexible data communications system implemented as an extension to, or as an alternative for, a wired LAN within a building or campus. Using electromagnetic waves, WLANs transmit and receive data over the air, minimizing the need for wired connections. Thus, WLANs combine data connectivity with user mobility, and, through simplified configuration, enable movable LANs. Some industries that have benefited from the productivity gains of using portable terminals (e.g., notebook computers) to transmit and receive real-time information are the digital home networking, health care, retail, manufacturing, and warehousing industries. 
     Manufacturers of WLANs have a range of transmission technologies to choose from when designing a WLAN. Some exemplary technologies are multicarrier systems, spread spectrum systems, narrowband systems, and infrared systems. Although each system has its own benefits and detriments, one particular type of multicarrier transmission system, orthogonal frequency division multiplexing (OFDM), has proven to be exceptionally useful for WLAN communications. 
     OFDM is a robust technique for efficiently transmitting data over a channel. The technique uses a plurality of sub-carrier frequencies (sub-carriers) within a channel bandwidth to transmit data. These sub-carriers are arranged for optimal bandwidth efficiency compared to conventional frequency division multiplexing (FDM) which can waste portions of the channel bandwidth in order to separate and isolate the sub-carrier frequency spectra and thereby avoid inter-carrier interference (ICI). By contrast, although the frequency spectra of OFDM sub-carriers overlap significantly within the OFDM channel bandwidth, OFDM nonetheless allows resolution and recovery of the information that has been modulated onto each sub-carrier. 
     The transmission of data through a channel via OFDM signals also provides several other advantages over more conventional transmission techniques. Some of these advantages are a tolerance to multipath delay spread and frequency selective fading, efficient spectrum usage, simplified sub-channel equalization, and good interference properties. 
     Although OFDM exhibits these advantages, conventional implementations of OFDM also exhibit several difficulties and practical limitations. One difficulty is the issue of determining and correcting for carrier frequency offset, a major aspect of OFDM synchronization. Ideally, the receive carrier frequency, f cr , should exactly match the transmit carrier frequency, f ct . If this condition is not met, however, the mis-match contributes to a non-zero carrier frequency offset, delta f c , in the received OFDM signal. OFDM signals are very susceptible to such carrier frequency offset which causes a loss of orthogonality between the OFDM sub-carriers and results in inter-carrier interference (ICI) and a severe increase in the bit error rate (BER) of the recovered data at the receiver. 
     Many OFDM standards require the transmission of pilots (known values) embedded in the user data. In conventional OFDM systems, it is common to average the pilots&#39; phase information to improve closed-loop carrier frequency offset tracking in a noisy environment. For example, the average of the pilots&#39; phases may be used to derive a carrier frequency offset estimation which, in turn, may be used to adjust the phase rotations of an equalizer&#39;s taps such that the effects of the carrier frequency offset are reduced or removed. One drawback to this technique is that, in the presence of a time-varying channel, the phases of the pilots may vary independently. More specifically, all the pilots&#39; phases share a common phase rotation representative of the carrier frequency offset caused by the mis-match between the transmitter carrier frequency and the receiver carrier frequency, as discussed above. However, in the presence of a time varying channel, each pilot phase may also contain an independent phase rotation caused by the transmission channel varying with time. These independent pilot phase rotations can potentially result in a destructive averaging of the pilots&#39; phases which, in turn, may corrupt the derivation of a carrier frequency offset estimation. A corrupted carrier frequency offset estimation may degrade the performance of any processing unit (e.g., an equalizer) that uses the estimation to compensate for the actual carrier frequency offset. The present invention is directed to the correction of this problem. 
     It is also possible that the frequency of the sampling clock of the receiver will differ slightly from the frequency of the sampling clock of the transmitter. If there is a frequency difference, the FFT window positioning with respect to the received signal can gradually drift over time. The time domain drift will result in a phase rotation of the received OFDM subcarriers in the frequency domain. The phase rotation may generate errors in the user data recovered by the OFDM receiver. The present invention is also directed to the correction of this problem. 
     BRIEF SUMMARY OF THE INVENTION 
     An Orthogonal Frequency Division Multiplexing (OFDM) receiver that employs N second-order phase-lock loops sharing a common integrator (where N is the number of pilots in the system). The N second order phase-lock loops track out independent pilot phase rotations to facilitate the constructive averaging of the pilots&#39; phase information. At the same time, by sharing a common integrator, the OFDM receiver takes advantage of noise averaging over multiple pilots to obtain a cleaner frequency offset estimation. The OFDM receiver may also compensate for FFT window drift by calculating a phase difference between a selected pair of pilots and tracking the rate of change of the calculated phase difference over time. The calculated phase difference is used to control the position of an upstream FFT window after a predetermined phase difference threshold is exceeded. The tracked rate of change is used to continuously adjust the phase of downstream equalizer taps. 
    
    
     
       BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS 
       In the drawings: 
         FIG. 1  is a block diagram of an exemplary OFDM receiver; 
         FIG. 2  is a diagram illustrating the placement of a training sequence, user data, and pilot signals within an OFDM symbol frame according to the present invention; 
         FIG. 3  is a block diagram illustrating a carrier frequency offset compensation system for an OFDM receiver according to the present invention; 
         FIG. 4  is a block diagram illustrating the present invention as integrated with the exemplary OFDM receiver of  FIG. 1 ; and 
         FIGS. 5 and 6  are graphical representations of common, independent, and adjusted phase errors. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     The characteristics and advantages of the present invention will become more apparent from the following description, given by way of example. 
     Referring to  FIG. 1 , the first element of a typical OFDM receiver  10  is an RF receiver  12 . Many variations of RF receiver  12  exist and are well known in the art, but typically, RF receiver  12  includes an antenna  14 , a low noise amplifier (LNA)  16 , an RF bandpass filter  18 , an automatic gain control (AGC) circuit  20 , an RF mixer  22 , an RF carrier frequency local oscillator  24 , and an IF bandpass filter  26 . 
     Through antenna  14 , RF receiver  12  couples in the RF OFDM-modulated carrier after it passes through the channel. Then, by mixing it with a receiver carrier of frequency fcr generated by RF local oscillator  24 , RF receiver  12  downconverts the RF OFDM-modulated carrier to obtain a received IF OFDM signal. The frequency difference between the receiver carrier and the transmitter carrier contributes to the carrier frequency offset, delta f c . 
     This received IF OFDM signal is coupled to mixer  28  and mixer  30  to be mixed with an in-phase IF signal and a 90° phase-shifted (quadrature) IF signal, respectively, to produce in-phase and quadrature OFDM signals, respectively. 
     The in-phase IF signal that feeds into mixer  28  is produced by an IF local oscillator  32 . The 90° phase-shifted IF signal that feeds into mixer  30  is derived from the in-phase IF signal of IF local oscillator  32  by passing the in-phase IF signal through a 90° phase shifter  34  before providing it to mixer  30 . 
     The in-phase and quadrature OFDM signals then pass into analog-to-digital converters (ADCs)  36  and  38 , respectively, where they are digitized at a sampling rate f ck     —     r  as determined by a clock circuit  40 . ADCs  36  and  38  produce digital samples that form an in-phase and a quadrature discrete-time OFDM signal, respectively. The difference between the sampling rates of the receiver and that of the transmitter is the sampling rate offset, delta f ck =f ck     —     r −f ck     —     t . 
     The unfiltered in-phase and quadrature discrete-time OFDM signals from ADCs  36  and  38  then pass through digital low-pass filters  42  and  44 , respectively. The output of lowpass digital filters  42  and  44  are filtered in-phase and quadrature samples, respectively, of the received OFDM signal. In this way, the received OFDM signal is converted into in-phase (qi) and quadrature (pi) samples that represent the real and imaginary-valued components, respectively, of the complex-valued OFDM signal, r i =q i +jp i . These in-phase and quadrature (real-valued and imaginary-valued) samples of the received OFDM signal are then delivered to FFT  46 . Note that in some conventional implementations of receiver  10 , the analog-to-digital conversion is done before the IF mixing process. In such an implementation, the mixing process involves the use of digital mixers and a digital frequency synthesizer. Also note that in many conventional implementations of receiver  10 , the digital-to-analog conversion is performed after the filtering. 
     FFT  46  performs the Fast Fourier Transform (FFT) of the received OFDM signal in order to recover the sequences of frequency-domain sub-symbols that were used to modulate the sub-carriers during each OFDM symbol interval. FFT  46  then delivers these sequences of sub-symbols to a decoder  48 . 
     Decoder  48  recovers the transmitted data bits from the sequences of frequency-domain sub-symbols that are delivered to it from FFT  46 . This recovery is performed by decoding the frequency-domain sub-symbols to obtain a stream of data bits which should ideally match the stream of data bits that were fed into the OFOM transmitter. This decoding process can include soft Viterbi decoding and/or Reed-Solomon decoding, for example, to recover the data from the block and/or convolutionally encoded sub-symbols. 
     Turning to  FIG. 2 , an exemplary OFDM symbol frame  50  of the present invention is shown. Symbol frame  50  includes a training sequence or symbol  52  containing known transmission values for each subcarrier in the OFDM carrier, and a predetermined number of cyclic prefix  54  and user data  56  pairs. For example, the proposed ETSI-BRAN HIPERLAN/2 (Europe) and IEEE 802.11a (USA) wireless LAN standards, herein incorporated by reference, assign 64 known values or subsymbols (i.e., 52 non-zero values and 12 zero values) to selected training symbols of a training sequence (e.g., “training symbol C” of the proposed ETSI standard and “long OFDM training symbol” of the proposed IEEE standard). User data  56  has a predetermined number of pilots  58 , also containing known transmission values, embedded on predetermined subcarriers. For example, the proposed ETSI and IEEE standards have four pilots located at bins or subcarriers ±7 and ±21. Although the present invention is described as operating in a receiver that conforms to the proposed ETSI-BRAN HIPERLAN/2 (Europe) and IEEE 802.11a (USA) wireless LAN standards, it is considered within the skill of one skilled in the art to implement the teachings of the present invention in other OFDM systems. 
     Referring now to  FIG. 3 , an exemplary embodiment of the present invention is shown. Although the present invention is illustrated as being distinct from the elements of OFDM receiver of  FIG. 1 , one skilled in the art will readily devise that the present invention may be integrated with the elements of the OFDM receiver, as shown in  FIG. 4  and discussed below. However, the present invention is illustrated as a distinct carrier frequency offset compensation system for clarity, ease of reference, and to facilitate an understanding of the present invention. 
     Referring now to  FIG. 3 , a carrier frequency offset compensation system  60  is shown. It should be noted that system  60  may be embodied in software, hardware, or some combination thereof. System  60  includes a plurality of second-order phase-lock loops that share a common integrator. As discussed below in further detail, the plurality of phase-lock loops enable the removal of independent pilot phase errors (i.e., phase rotations) caused by a time varying channel and, thereby, facilitates the constructive averaging of the pilots&#39; phase information for deriving a carrier frequency offset estimate. It should be further noted that by sharing a common integrator, noise averaging is advantageously taken over multiple pilots to derive a cleaner carrier frequency offset estimate. 
     More specifically, there are N second-order phase-lock loops (PLLs) where N represents the number of pilots processed by system  60 . Each second-order PLL includes a derotator or complex multiplier  62 , a phase error detector  64 , a proportional gain stage  66 , a summer  74 , a numerically controlled oscillator (NCO)  76 , and a Sin/Cos look-up table  78 . The second-order PLLs also share an averaging unit  68 , an integral gain stage  70  and an integrator  72  that are coupled between phase detector  64  and summer  74  of each PLL. A Sin/Cos table  80  may be coupled to the output of integrator  72  and to an input of an equalizer  82 . Furthermore, a phase difference calculator  84 , comparator  86  and FFT window offset corrector  88  arrangement may be coupled to the output of the NCOs  76  of the second order PLLs, as discussed in further detail below. 
     In operation, each pilot  58  of a user data segment  56  is processed by a separate PLL and is averaged with the other pilots  58  of a user data segment  56 . More specifically, each derotator  62  multiplies a received pilot with a complex number (representing an independent phase error correction) to drive the independent phase error towards zero. 
     Each derotator  62  passes the processed pilot to a phase error detector  64 . Each phase error detector  64  derives a phase error of the pilot. One exemplary way to derive phase error is by calculating the difference between a known ideal phase of the pilot and the actual phase of the received pilot. The use of other phase error derivation techniques, as known by those skilled in the art, is considered within the scope of the present invention. In each PLL, the phase error is passed to an associated proportional gain stage  66  as well as to the shared averaging unit  68 . Each proportional gain stage  66  scales the received phase error (representing the independent phase rotation of the pilot) to a predetermined increment usable by the associated NCO  76  of each PLL. Averaging unit  68  averages the phase error values received for all the pilots in a given user data segment and passes the average error (representing the average phase rotation for all of the pilots in a given user segment) to integral gain stage  70 . Integral gain stage  70  scales the average phase error to a predetermined increment usable by each NCO  76  as well as by Sin/Cos lookup table  80 , as discussed in further detail below. Integrator  72  integrates the scaled average phase errors received from integral gain stage  70  and outputs an integrated scaled phase error representing the common phase error for all the pilots over multiple user data segments. It should be noted a portion of the integrated scaled phase error will be due to the independent phase rotations of the pilots until a certain lock condition is reached, as discussed below. 
     The summer  74  of each PLL sums the independent phase error received from the associated proportional gain stage  66  and the common phase error received from integrator  72 . The resulting value represents the common phase rotation for all the pilots as adjusted by the independent pilot phase rotation caused by the time varying channel. Referring now to  FIGS. 5 and 6 , graphical illustrations of common phase errors, independent phase errors, and adjusted phase errors are shown for two PLLs. 
     The adjusted phase error output from each summer  74  is passed to an associated NCO  76  that accumulates received phase errors over time. Each lookup table  78  of a given PLL converts the output of an associated NCO  76  into a phasor. The phasor is passed back to the associated derotator  62  which multiplies the next received pilot with the phasor to rotate the pilot such that the independent phase error is driven towards zero. 
     In addition to the second-order phase-lock loops, a Sin/Cos table  80  is coupled to integrator  72  for converting, after a predetermined lock condition, the output of integrator  72  into a carrier frequency-offset estimate used by equalizer  82  to adjust the phase rotation of the equalizer taps. It should be noted that the carrier frequency-offset estimate may be passed to other processing units (not shown) for correction of the carrier frequency offset. One exemplary lock condition is when the output of integrator  72  falls within a predetermined range over a predetermined period of time. Such an occurrence indicates that the PLLs have reached steady state and the independent phase errors have been removed. It should be noted that a single time-shared PLL may preferably be used instead of the plurality of PLLs shown in  FIG. 3  to process the pilots and remove the independent phase errors from the pilots. 
     As discussed above, the frequency of the sampling clock of the receiver may differ slightly from the frequency of the sampling clock of the transmitter. If there is a frequency difference, the FFT window positioning with respect to the received signal can gradually drift over time. The FFT window drift will result in a phase rotation of the received OFDM subcarriers. The phase rotation may generate errors in the user data recovered by the OFDM receiver. The phase difference calculator  84 , rate of phase difference estimator  86  and FFT window synch unit  88  arrangement is directed to compensation and correction of the FFT window offset. 
     More specifically, phase difference calculator  84  calculates the difference between the values output from a given pair of NCOs  76 . This difference is equal to the phase difference between a given pair of pilot subcarriers in a user data segment. It should be noted that the NCO values will roll over after reaching ±pi. Therefore, phase difference calculator  84  tracks the number of times the value of each NCO  76  exceeds ±pi to accurately calculate the phase difference between the values output from a given pair of NCOs  76 . The calculated phase difference is passed to rate of phase difference estimator  86  and FFT window synch unit  88 . FFT window synch unit  88  compares the calculated phase difference to a threshold phase difference (e.g., a phase difference representative of an FFT window offset of one sample) and controls the position of an upstream FFT window (e.g., shifts the window by a sample) if the calculated phase difference exceeds the threshold phase difference. In this manner, the FFT window offset of an upstream FFT may be periodically corrected when the calculated phase difference exceeds a predetermined threshold phase difference. Estimator  86  tracks the rate of change of the calculated phase difference over multiple user data segments. Estimator  86  generates an equalizer adjustment value that can be combined (e.g., via summer  74 ) with the output of integrator  72  such that the carrier-frequency offset output from Sin/Cos table  80  is compensated for the continuously changing phase difference between the values output from a given pair of NCOs  76 . In this manner, the equalizer taps of a downstream equalizer can be continuously rotated to compensate for a gradually drifting FFT window. 
     Referring now to  FIG. 4 , the present invention is integrated with the exemplary OFDM receiver of  FIG. 1 , as shown. More specifically, system  60  is coupled to the outputs of FFT  46  and to the inputs of a processing unit that compensates for a carrier frequency error (e.g., equalizer  82  of  FIG. 3  and/or a front end frequency offset correction unit (not shown)). Furthermore, an output of system  60  is fed back to FFT  46 . With this arrangement, system  60  extracts pilots from the OFDM samples received from FFT  46  and derives a frequency offset estimate free of the independent phase errors caused by a time varying channel. System  60  also processes the extracted pilots to compensate (e.g., in equalizer  82 ) and correct (e.g., in FFT  46 ) for FFT window drift.