Abstract:
A control circuit and method for controlling a sensorless brushless DC motor are provided which are capable of effectively filtering noise. The control circuit includes: a phase voltage measuring unit for determining the voltage of each phase from the terminal voltages of the stator coils; a phase voltage sign detecting unit for detecting the sign of the phase voltage measured from the phase voltage measuring unit and outputting a phase voltage sign signal; and a digital phase-shifter for generating a delay signal which is delayed up to 90° with respect to the phase voltage sign signal and supplying the delayed signal to the commutation unit.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to a sensorless brushless DC motor (hereinafter referred to as sensorless BLDC motor) used where the temperature or humidity is high or a position sensor is difficult to establish, and more particularly, to an apparatus and method for controlling a sensorless BLDC motor which effectively filters out the influence of noise. 
     2. Related Art 
     The electronic industry has been investigating brushless DC motors because of the small size of such motors. In a brushless motor, it is essential to exactly measure the position of a motor rotor to exactly supply the voltage and a position measuring apparatus can be used for this purpose. However, when a sensor of the rotor position measuring apparatus does not operate properly due to high humidity or temperature, or, in an apparatus, such as a compressor, in which it is impossible to employ such a position measuring apparatus, it becomes difficult to exactly measure the rotor position. Moreover, because the space occupied by the rotor position measuring apparatus in the BLDC motor is so large, a sensorless BLDC motor is desirable. 
     Such a sensorless BLDC motor controls the power supplied to a stator coil based upon an induced voltage at the stator coil terminal. In a sensorless BLDC motor, the rotor has no position sensor. 
     An edge detection signal is generated by detecting the edge of the waveform of a comparative output which is obtained by comparing an induced voltage generated at each terminal of a plurality of stator coils located along with the rotational direction of the stator with a standard electric potential (0 V). 
     Presently, in order to remove a variation in the output caused by a spike voltage Vs produced when the power supply is turned on or off, the compared output is standardized. This standardized compared output is used to control power supplied to each stator coil, i.e., commutation is performed based on this output. 
     However, in instances where the edge signal is generated at a point at which the power of the stator coils is not turned on or off, the system will not operate properly. In other words, when commutation time of the stator coil is inaccurate, the motor is not rotated smoothly. 
     In order to solve the problem discussed above, improved motors have been developed. The motor disclosed in Japanese Patent Laid-open No. 4-317585 (1992), which is one of the improved motors, will now be considered. 
     FIG. 1 is a block diagram illustrating a control circuit of a conventional sensorless BLDC motor while FIG. 2 is a waveform illustrating the operating states of the units of FIG. 1. Referring to FIGS. 1 and 2, stator coils Lu, Lv, and Lw are arranged according to the rotational direction of the rotor, and are connected to a commutation signal generating circuit 1. The control circuit of FIG. 1 also includes a multiple phase driving circuit 2, a zero level comparison circuit 3, and an edge detection circuit 4. In addition, the control circuit includes a timing control circuit 5, an edge detection control circuit 6, and a starting oscillation circuit 8 as well as a clock converting or switching circuit 8 and a control unit 9. 
     The commutation signal generating circuit 1 generates the multiple phase control signal for controlling the power supply to each stator coil Lu, Lv and Lw, i.e., the commutation signals, according to a signal clock CK. The multiple phase driving circuit 2 supplies the power to drive stator coils Lu, Lv and Lw according to the phase based upon the commutation signals. The zero level comparison circuit 3 compares the induced voltages U-N, V-N and W-N generated at each terminal of the stator coils Lu, Lv and Lw with a standard electrical potential 0 V. A hysteresis comparison circuit having a predetermined input value is preferably used as the zero level comparison circuit 3. The edge detection circuit 4 detects the edge of the output wave forms of the comparison circuit 3 according to their phase, and logically adds an edge detection pulse of each phase, thereby generating a standard clock signal CK2 of the commutation signal generating circuit 1. The timing control circuit 5 delays the standard clock signal CK2 generated by the edge detection circuit 4 as much as a predetermined time t2, and transmits a delayed standard clock signal CK used in generating the commutation signals of the commutation signal generating circuit 1. 
     The edge detection control circuit 6, which includes a delay circuit 61 and an OR logic gate 62, prevents the edge detection of the waveforms of the compared outputs of each corresponding phase when the power supplied to the stator coils Lu, Lv and Lw is turned on or off, based upon a status signal indicating the operating state of the commutation signal generating circuit 1. The oscillation circuit 7 generates a standard clock signal CK1 when the motor starts. At this time, the switching or converting circuit 8 receives the standard clock signal CK1 from the oscillation circuit 7, and transmits the standard clock signal CK to the commutation signal generating circuit 1 through the timing control circuit 5. After the motor is energized, i.e., driven, the switching circuit 8 receives the standard clock signal CK2 from the edge detection circuit 4, and transmits the standard clock signal CK to the commutation signal generating circuit 1 through the timing control circuit 5. The control unit 9 controls the switching circuit 8 according to the standard clock CK2 outputted from the edge detection circuit 4. 
     Considering the operation of the control circuit of FIG. 1, the driving circuit of the sensorless BLDC motor derives the standard clock signal CK used for controlling commutation by comparing the induced voltages U-N, V-N and W-N generated at each terminal of the stator coils Lu, Lv, and Lw with each standard electrical potential 0 V, and by detecting each edge of the waveforms of the compared outputs according to their phases. At this time, the influence of the spike voltage Vs generated by switching of the power supply of each stator coil Lu, Lv, and Lw is detected by the comparison circuit 3. Before being detected by the edge detection circuit 4, the spike is selectively precluded from detection by the edge detection control circuit 6. However, depending upon the control state, when the phase voltage is measured in a transition state or when there is a heavy motor load, noise that interferes with edge detection can be produced even at points where the power supplied to the stator coils is not turned on or off. Moreover, the prior art motors allow the induced voltages to exceed the zero voltage level while the edge detection is prohibited and disturbs the precision of the commutation signals. 
     SUMMARY OF THE INVENTION 
     In accordance with the invention, a control circuit and method are provided for controlling the magnetization of a plurality of stator coils by a power supply signal supplied from a commutation unit to a sensorless brushless direct current (BLDC) motor. 
     According to one aspect of the invention, the control circuit includes: a phase voltage determining means for determining the voltage of each phase from the terminal voltages of the stator coils; a phase voltage sign detecting means for detecting the sign of the phase voltage measured from the phase voltage determining means and outputting a phase voltage sign signal; and a digital phase-shifter for generating a delay signal which is delayed up to 90° with respect to the phase voltage sign signal and supplying the delayed signal to the commutation unit. 
     In a preferred embodiment, the digital phase-shifter of the control circuit includes: a first counter for counting to a higher value when the phase voltage sign signal has a high state; a second counter for counting to a higher value when the phase voltage sign signal has a low state; a comparing means for determining whether the value of one of the counters is at least equal to 1/2 the value of the other of the counters and less than the value of the other of the counters; a clear means for resetting the other of the counters to a value of zero when the value of the one of the counters is at least equal to 1/2 the value of the other of the counters and less than the value of the other of the counters; and a delay signal generating means for generating a first signal when the first counter is reset to zero by the clear means, and generating a second signal of a different state from the first signal when the second counter is reset to zero by the clear means. 
     In an even more preferred embodiment, the digital phase-shifter of the control circuit further includes: a set value comparison means for comparing the value of each of the counters with a predetermined maximum value; and a set value maintaining means for maintaining the value of a compared counter at the maximum value when the value of the compared counter is not less than said maximum value. 
     In another aspect of the invention, the control includes: a stepping signal generating means for generating stepping signals for starting the motor; a phase voltage detecting means for detecting the voltages of each phase of stator coils of the motor; a phase voltage sign detecting means for detecting the polarity of each of the phase voltages detected by the phase voltage detecting means and outputting corresponding phase voltage sign signals; a digital phase-shifter for outputting phase shifted signals delayed up to 90° with respect to the phase voltage sign signals; a mode selection means for comparing the stepping signals with the phase shifted signals and outputting the stepping signals until the motor reaches a predetermined speed and outputting the phase shifted signals after the motor reaches the predetermined speed; a commutation unit for outputting the signal selected by the mode selection unit as a power supply signal of the stator coils; and an inverter for supplying the electric power to the stator coils of the motor according to the power supply signal of the commutation means. 
     In a preferred embodiment, the digital phase-shifter comprises three phase shifter units, each of the phase shifter units includes: a first counter for counting to a higher value when the phase voltage sign signal is a logic high signal; a second counter for counting to a higher value when the phase voltage sign signal is a logic low signal; a comparator for determining if the value of one counter is at least equal to the value of the other of the counters and less than the value of the other of the counters; and a delay signal generating means for changing a delay signal which changes when the value of one of the counters is at least equal to the value of the other of the counters and less than the value of the other of the counters. 
     In accordance with another aspect of the invention, a control method motor for controlling the magnetization of a plurality of stator coils by a power supply signal applied by a commutation unit is provided for a sensorless brushless DC motor. The method includes the steps of: determining each phase voltage from the terminal voltages of each phase of the stators; detecting the sign of the phase voltages measured by the step of determining the phase voltages; and a digital phase-shifting for generating signals delayed up to 90° with respect to the phase voltage signals from the step of detecting the sign of the phase voltage, and outputting the delayed signals to the commutation unit. 
     In a preferred embodiment, the step of digital phase-shifting includes the steps of: counting to a higher value with a first counter when the sign of the phase voltage detected by said step of detecting the sign of the phase voltage is positive; counting to a higher value with a second counter when the sign of the phase voltage detected by the step of detecting the sign of the phase voltage is positive; comparing the values in the first and second counters for determining whether the value of one of the counters is greater than at least 1/2 the value of the other of the counters and less than the value of the other of the counters; resetting the other of the counters to a value to zero when the one of the counters is at least equal to one-half the value of the other of the counters and less than the value of the other of the counters; generating a delay signal for generating a logic low signal when the first counter is reset in the resetting step and generating a logic high signal when the second counter is reset in the resetting step. 
     In an even more preferred embodiment, the step of digital phase-shifting further includes the steps of: comparing the value of each of the counters with a predetermined maximum value; and maintaining the value of any of the counters at the maximum value when the value of the respective counter is not less than the maximum value. 
     Other features and advantages of the invention will be set forth in, or apparent from, the following detailed description of preferred embodiments of the invention. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The above object and advantages of the present invention will be more clearly understood from the following detailed description thereof, taken in conjunction with the accompanying drawings, in which: 
     FIG. 1 is a block diagram illustrating a control circuit of a conventional sensorless BLDC motor, as described above; 
     FIG. 2 is a waveform diagram illustrating the operating state of key units of the control circuit shown in FIG. 1; 
     FIG. 3 is a block diagram of a control circuit in accordance with a first embodiment of the present invention; 
     FIG. 4 is a diagram illustrating the waveforms of key units of the control circuit shown in FIG. 3; 
     FIG. 5 is a circuit diagram of a preferred embodiment of the digital phase-shifter of FIG. 3; 
     FIG. 6 is a flow chart illustrating the method of forming a signal delayed as 90° using a digital phase-shifter; and 
     FIG. 7 is a waveform diagram illustrating signals H1, H2 and H3 and driving signals of the inverter of FIG. 3. 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Referring to FIG. 3, a starting signal generating unit 110, indicated as being part of a microcomputer 100, generates stepping signals H1&#39;, H2&#39; and H3&#39; until a motor 170 arrives at a prescribed speed after initial starting of the motor 170. A phase voltage detecting unit 120 calculates the phase voltage from each terminal voltage Va, Vb and Vc of the three phases of motor 170 and detects the signs of each phase voltage. A digital phase-shifter 130 generates a signal delayed up to 90° with respect to the input waveform of the phase voltage detected by the phase voltage detecting unit 120. A mode selection unit 140, connected to starting signal generating unit 110 and to digital phase shifter 130, determines whether the motor 170 arrived at a prescribed speed in the starting stage of the motor. A commutation unit 150 connected to the output of mode selection unit 140 provides a power supply signal in accordance with the signal received from the digital phase-shifter 130 or the signal from the starting signal generating unit 110 depending on the state of mode selection unit 140. An inverter 160 accelerates the motor 170 according to the speed information obtained from the digital phase-shifter 130 and a voltage command of pulse width modulation (PWM) form (not shown). The sensorless BLDC motor 170 is operated by the command, i.e., based on the output of the inverter 160. In addition, the functions of the mode selection unit 140, the commutation unit 150, and the digital phase-shifter 130 can be performed by microcomputer 100 as indicated above for starting signal generator 110. 
     The phase voltage detecting unit 120 measures voltages between the stator terminals Va, Vb and Vc of motor 170 which are obtained from the following equations: 
     
         Vab=Va-Vb, Vbc=Vb-Vc, and Vca=Vc-Va 
    
     The voltage Vn of the central point is obtained from the phase voltages shown in the following equations by using the equation, Vn=(Va+Vb+Vc)/3. 
     
         Van=Va-Vn=(2Va-Vb-Vc)/3=(2Vab+Vbc)/3 
    
     
         Vbn=Vb-Vn=(2Vb-Vc-Va)/3=(2Vbc+Vab)/3, and 
    
     
         Vcn=Vc-Vn=-Van-Vbn 
    
     As described in more detail below in connection with FIG. 5, the phase voltage signs Sa, Sb and Sc are obtained by using a hysteresis comparator after decreasing the PWM frequency component from the phase voltages Van, Vbn and Vcn by using a low pass filter (LPF). 
     As shown in FIG. 4, the phase voltage Van has a waveform which is processed to perform the above-mentioned LPF function. The phase voltage passes through the hysteresis comparator mentioned above and a phase voltage sign indicated as Sa, is determined. At this time, the phase voltage sign signal Sa has edge portions, i.e., leading and trailing edges, where the induced voltage Van passes through the zero voltage level. 
     The digital phase-shifter 130 records the phase voltage sign which is detected by the phase voltage detecting unit 120 during each sampling period through the use of a positive counter and a negative counter described in more detail below in connection with FIG. 5. When the value of a first counter arrives at half the value of the second counter, the second counter is cleared. The two counters are cleared alternately. According to this operation, the output is reset and there is generated the output signal mentioned above which is delayed as much as 90° with respect to the basic waveform. 
     FIG. 5 is a simplified circuit diagram of a preferred embodiment of the digital phase-shifter 130 of FIG. 3 showing the basic elements thereof. As shown, the digital phase-shifter 130 includes a first digital phase-shifter unit 130a, a second digital phase-shifter unit 130b and a third digital phase-shifter unit 130c each of which provides a signal phase-shift by receiving respective phase voltage signs Sa, Sb or Sc from the phase voltage detecting unit 120. Because the digital phase-shifter units 130a, 130b and 130c use the same processing operation for providing a signal phase-shift excluding the input/output, only the first digital phase-shifter unit 130a would be considered in detail. 
     Referring to unit 130a, a positive counter 131a (hereinafter referred to as the Pa counter) performs a counting operation when the phase voltage signal Sa of phase A is at a logic &#34;high&#34; state. In contrast, a negative counter 132a (hereinafter referred to as the Na counter) performs a counting operation when the phase voltage signal Sa of phase A is at a logic &#34;low&#34; state. 
     The value or count of the Na counter 132 is divided by 2 by a first divider 133a. A positive sign comparator 134a compares the value of the Pa counter 131a with 1/2 the value of the Na counter 131a. When the value or count of the Pa counter 132a is first equal to or greater than the output value of the first divider 133a, i.e., greater than 1/2 the value of counter 132a, the value of the Na counter 132a is set to zero and the low state of the output of unit 130a is converted into the high state. This output state is maintained until the value of the Pa counter 131a drops below the output value of the first divider 133a or counter 131a is cleared by the value of the Na counter 132a. 
     The value of the Pa counter 131a is divided by 2 by a second divider 135a and a negative sign comparator 136a compares the value of the Na counter 132a with 1/2 the value of the Pa counter 131a. As a result of the comparison, when the value of the Na counter 132a is first equal to or greater than the output value of the second divider 135a, the value of the Pa counter 131a is set to zero as shown in FIG. 4. 
     FIG. 6 is a flow chart illustrating the method or process for forming the aforementioned signal delayed as much as 90° by the digital phase-shifter 130. As shown, a determination is first made as to whether the phase voltage Sa of phase `A` is 1 (S1). In the event that the phase voltage Sa is 1, the count of a counter Pa having an initial count or value of 0 is increased by 1 block (S2), while in the event that the phase voltage Sa is not 1, the count of a counter Na having an initial count or value of 0 is increased by 1 block (S3). Next, a determination is made whether the count or values of each of the counters Pa and Na is smaller than a predetermined value M decision diamonds (S4 and S5). The predetermined value M is a very large value and prevents overflow of the counts or values of each counter. When determining step indicated by decision diamond S4, determines that the value of the counter Pa is smaller than the value M, a determination is then made as to whether the value of the counter Pa corresponds to one-half the value of the counter Na block (S7). On the other hand, when the value of the counter Pa is larger than the value M, the value of the counter Pa is set as the value M block (S6) and a determination is made as to whether the value of the counter Pa corresponds to one-half the value of the counter Na decision diamond (S7). When, in the step represented at S7, the value of the counter Pa is equal to one-half the value of the counter Na, the counter Na is initialized, i.e., reset to zero, and a phase-shifted output Sao is set to 1. On the other hand, wherein the step represented at S7, the value of the counter Pa is not one-half the value of the counter Na, the procedure ends for the present time period. 
     As shown, in the step represented by decision diamond S5, it is determined whether the value of the counter Na is smaller than the predetermined value M. When the value of the counter Na is smaller than the value M, it is determined whether the value of the counter Na is at least one-half the value of the counter Pa decision diamond (S8). When the value of the counter Na is not smaller than the value M, the value of the counter Na is set as the value M and a determination is made as to whether the value of the counter Na (Na =M) is at least one-half the value of the counter Pa decision diamond (S8). When, in the determining step indicated at S8, the value of the counter Na is at least one-half the value of the counter Pa, the counter Pa is initialized, i.e., reset to zero and the phase-shifted output Sao is reset to zero. On the other hand, when, in the determining step indicated at S8, the value of the counter Na is not at least one-half the value of the counter Pa, the procedure ends for the present time period. 
     The operation of the control circuit of FIG. 3 and the effects of noise on the circuit are illustrated in FIG. 4. For ease of explanation, only a single phase will be described. The terminal current of the &#34;A&#34; terminal Ia is shown at the top of the figure. As illustrated, the phase voltage Van is not a perfect sine wave. The rapid changes in the terminal current Ia cause the phase voltage Van to have voltage spikes. To precisely and accurately form the commutation signal H1, these voltage spikes must be filtered out. 
     The phase voltage detecting unit 120 of FIG. 4 determines the polarity of the phase voltage Van. As shown in FIG. 4, when Van is positive, the detecting unit 120 outputs a logic high signal or logical &#34;1&#34; which is the state of the phase voltage sign signal Sa. On the other hand, when phase voltage Van is negative, the phase voltage sign signal Sa is at a logic &#34;low&#34; state or a logical &#34;0&#34;. The count of either counter Pa 131a or counter Na 132a in unit 130a of the digital phase shifter 130 is increased based on the logic level of phase voltage sign signal Sa. 
     Referring again to FIG. 4, the phase voltage Van is shown as being positive from t0 to t5. As a result, the phase voltage sign signal Sa has a logic high value. While Sa is in a high state, the valve in counter Pa 131a increases in steps. FIG. 4 shows the values in counters Pa 131a and Na 132a increasing continuously for illustration purposes only. The values in the counters actually increase in discrete steps. The sampling period of the counters is chosen such that several samples are taken for each time interval shown in FIG. 4. In one preferred embodiment, the sampling period is 256 μsec and the counters Pa 131a and Na 132a increase in steps of 1/(256 μsec×15 V). It is important to note that counter Pa 131a has a count or value greater than 0 in the time period immediately before t0, for reasons that will be explained below. As shown, at t3 the value in counter Pa 131a is greater than to the value of counter Na 132a. Since Pa exceeds 1/2 the value of Na, counter Na is reset to 0 and phase-shifted output Sao becomes &#34;1&#34; or high. 
     At t5, a voltage spike Vsn occurs. From t5 to t5&#39; Van is negative. A negative Van results in phase voltage sign signal Sa assuming a logic low value and the count of counter Na 132a increasing. From t5&#39; to t6, phase voltage Van returns to a positive value. Therefore, phase voltage sign signal Sa returns to a logic high state and the count or value in counter Pa 131a increases. Since the voltage spike occurs within a single sampling period, the value in counter Pa 131a and the value in counter Na 132a both increase. 
     At the end of t5, the value in counter Pa 131a exceeds 1/2 of the value in counter Na 132a. However, as illustrated, counter Na 132a is not reset after t5 because each counter is only reset when the value or count thereof is greater than 1/2 the value of the other counter and a value less than the value of the other counter. 
     From t6 to t11, the phase voltage Van is negative and the voltage sign signal Sa has a logic low value. As stated above, when the voltage sign signal Sa is at a logic low level, the count or value of counter Na 132a increases. The phase voltage Van crosses the 0 axis at t6. Counter Na 132a has a value greater than 0 immediately before t6, as a result of voltage spike Vsn. At t8, the value in counter Na 132a exceeds 1/2 the value in counter Pa 131a. Since the value in counter Na 132a exceeds 1/2 the value in counter Pa 131a, counter Pa is reset to 0 and the phase-shifted output Sao becomes a logical 0 at t9. Accordingly, the polarity of the phase shifted output Sao matches the polarity of the phase voltage Van shifted by 90° and without the spikes. 
     The circuit of FIG. 3 continues to operate in a similar fashion to that described above. For every 360° cycle of the motor 170, there is one negative voltage spike Vsn in the positive portion of the phase voltage Van and one positive voltage spike Vsp in the negative portion of the phase voltage Van. The voltage spikes are effectively filtered out by increasing the values in both counters Pa 131a and Na 132a during the voltage spikes. The phase shifted output Sao is, as a result, a function of the difference between the two counters. Using the above-described method, the waveforms of the phase-shifted outputs Sbo Sco of each phase can be obtained. Commutation signals H1, H2 and H3 are supplied by inverting the signals Sao, Sbo and Sco. 
     FIG. 7 is a waveform diagram illustrating signals H1, H2 and H3 as well as the driving signals A+, B+, C+, A-, B- and C- of the inverter 160. The commutation signals H1, H2 and H3 correspond to inverted versions of phase-shifted signals Sbo, Sco and Sao. At this time, the commutation signals H1 and H2 have a phase difference of 120°, and similarly, H2 and H3 have a phase difference of 120°. The waveforms A+, B+, C+, A-, B- and C- shown in FIG. 7 are supplied to a switching element of the inverter 160 and turn on the power supply. The phase voltages Van, Vbn and Vcn are shown without the voltage spikes to more clearly show the phase differences. 
     As described above, the digital phase-shifter integrates the measured phase voltage signals and, in effect, filters out the influence of noise when measuring the phase voltages. Accordingly, more stable and exact commutation signals can be obtained in the presence of overload or any external noises. Moreover, in the case of a heavy load, this system makes the current transition time lead the commutation time, and compensates for the current response lagging, thereby increasing the efficiency of the system. 
     Although the preferred embodiments of the present invention have been disclosed for illustrative purposes, those skilled in the art will appreciate that various modifications, additions and substitutions are possible without departing from the scope and spirit of the invention as defined in the accompanying claims.