Abstract:
An unregulated inductorless direct current to direct current converter comprising a first voltage-to-current converter configured to convert a first voltage to a first current and a second voltage-to-current converter configured to convert a second voltage to a second current. A regulation circuit is coupled to the first and second voltage-to-current converters and configured to generate an output current proportional to the difference between the first and second currents. Also a variable frequency oscillator is coupled to the regulation circuit, the oscillator receiving as a control current the output current therefrom and outputting a clock signal having a frequency proportionate to the control current. The converter further comprises an output stage coupled to receive the clock signal and receiving an input voltage and outputting an output voltage, the output voltage and the input voltage having a ratio that is determined by the clock signal.

Description:
TECHNICAL FIELD OF THE INVENTION 
     This invention relates to DC/DC converters and more particularly to a frequency control circuit for regulation of an inductorless DC/DC converter. 
     BACKGROUND OF THE INVENTION 
     Unregulated inductorless DC/DC converters (i.e. charge pumps) are used to double, triple, or invert a voltage that is supplied to the converter. These converters, however, do not generate a constant output voltage. An exemplary unregulated converter comprises a clock generator or oscillator and an array of power switches. FIG. 1A, for example, shows a prior art unregulated voltage doubler  10 . The voltage doubler  10  is coupled to a clock generator, as shown in FIG. 1B, which generates a signal A and B that serve as inputs into the voltage doubler  10 . These signals will turn the multiple transistors M 1 , M 2 , M 3 , and M 4  on as needed to double the voltage V in . When transistors M 2  and M 1  are turned on (i.e. during phase A when the signal is high), the capacitor C f  will be charged to the voltage V in . When transistors M 3  and M 4  are turned on (i.e. during phase B when the signal is high), the capacitor C f , which is already charged to the voltage V in , will be put in series to the input voltage, V in  and the output capacitor, C out  will be charged up to twice the voltage V in . 
     In order for the converter to work most efficiently and to be cost effective, there are four independent requirements that must be met. These requirements include 1) a small internal resistance (i.e. a low voltage drop at full load), 2) a low output voltage ripple at full load, 3) a low quiescent current, and 4) small and inexpensive external components (i.e. capacitors). The internal resistance R i2  of an inductorless transistor SC voltage doubler can be calculated as:                R   i2     =       1       C   f                *                f   clk         +                2              *       ∑     i   =   1     4          R   Mi                   (   1   )                                
     were C f =pump or “flying” capacitor, f clk =clock frequency, and R Mi =ON-resistance of switch M i . Thus, to minimize the internal resistance R i2 , a high clock frequency f clk  and/or a large flying capacitor C f  is needed. The output voltage ripple is represented by:                V   RIPPLE     =       1     2              *                C   OUT                *                f   clk         *                I   LOAD               (   2   )                                
     where V RIPPLE =ouput voltage ripple, C OUT =output capacitance, and I LOAD =load current. To minimize the output voltage ripple, then, a high clock frequency and/or a large output capacitance is needed. 
     Since the Power MOSFETS in the converter periodically have to change states, their gates periodically need to be charged and discharged. The gates of all power transistors in the converter can be seen as a capacitor which needs to be charged to the input voltage and discharged to ground. When a capacitor is charged from zero to any other voltage, half of the energy gets lost. Compared to an inductive converter, a Charge Pump has higher switching losses from the gate capacitances of the power transistors, since there are more transistors to control. These losses are represented by a quiescent current:                I   Q     =       (       V   IN     *                f   clk       )                *                  ∑     i   =   1     4                     C   Mi                 (   3   )                                
     To minimize the quiescent current, a low clock frequency is need. However, the low clock frequency needed to minimize the quiescent current is counter to the high clock frequency which is needed to minimize the internal resistance and the voltage ripple. Thus, it is impossible for the prior art to fulfill all four requirements because prior art devices run at a constant frequency. 
     In the prior art, designers have generally compromised on the conflicting performance characteristics and offered their devices with different operating frequencies, for example 1, 10, 50, and 100 khz. Thus, consumers must choose to fulfill certain of the requirements while forfeiting others. For example, consumers can typically obtain the first three requirements but at the expense of very costly, large external capacitors which allow a small operating frequency. If the quiescent current I Q  is not an issue, then high frequency versions with small external capacitors can be utilized. If internal resistance and voltage ripple is not an issue, low frequency versions with small external capacitors can be used, an example of which is the MAX 828 or a like device. Thus, what is needed is a design that will provide efficient and cost effective operation by meeting all four of the requirements. 
     SUMMARY OF THE INVENTION 
     These and other problems are generally solved or circumvented, and technical advantages are generally achieved, by the present invention that is a frequency control circuit for unregulated inductorless DC/DC converters. 
     In a preferred embodiment method of the present invention, an unregulated inductorless direct current to direct current converter comprises a first voltage-to-current converter configured to convert a first voltage to a first current and a second voltage-to-current converter configured to convert a second voltage to a second current. A regulation circuit is coupled to the first and second voltage-to-current converters and configured to generate an output current proportional to the difference between the first and second currents. The unregulated inductorless direct current to direct current converter further comprises a variable frequency oscillator coupled to the regulation circuit, the oscillator receiving as a control current the output current therefrom and outputting a clock signal having a frequency proportionate to the control current; and an output stage coupled to receive the clock signal and receiving an input voltage and outputting an output voltage, the output voltage and the input voltage having a ratio that is determined by the clock signal. 
     One advantage of a preferred embodiment of the present invention is that it provides a variable frequency to provide for small internal resistance, low output voltage ripple and quiescent current while allowing for small external capacitors. 
     Another advantage of a preferred embodiment of the present invention is that it defines a frequency sweep range that increases efficiency and decreases power loss. 
     Yet another advantage of a preferred embodiment of the present invention is that it provides for a cost effective device by allowing for the use of cheaper external components. 
     A further advantage of a preferred embodiment of the present invention is that it can flexibly operate with a variety of external components. 
     The foregoing has outlined rather broadly the features and technical advantages of the present invention in order that the detailed description of the invention that follows may be better understood. Additional features and advantages of the invention will be described hereinafter, which form the subject of the claims of the invention. It should be appreciated by those skilled in the art that the concepts and specific embodiments disclosed may be readily utilized as a basis for modifying or designing other structures or processes for carrying out the same purposes of the present invention. It should also be realized by those skilled in the art that such equivalent constructions do not depart from the spirit and scope of the invention as set forth in the appended claims. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     For a more complete understanding of the present invention, and the advantages thereof, reference is now made to the following descriptions taken in conjunction with the accompanying figures, in which: 
     FIGS. 1A-1B illustrate a prior art charge pump and its principle of operation; 
     FIG. 2 is a simple block diagram of the present invention; 
     FIG. 3 is a block diagram of a preferred embodiment of the present invention; and 
     FIG. 4 is a schematic of a preferred embodiment of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     The making and using of the presently preferred embodiment is discussed in detail below. It should be appreciated, however, that the present invention provides many applicable inventive concepts that can be embodied in a wide variety of specific contexts. The specific embodiments discussed are merely illustrative and do not limit the scope of the invention. 
     FIG. 2 is a simple block diagram of an inductorless DC/DC converter  20  implementing a frequency control circuit of the present invention. The converter  20  includes a frequency control circuit  22  coupled to a variable frequency oscillator (VFO)  24  to control the frequency of a charge pump output stage  26 . The frequency control circuit  22  measures an input voltage V in  and output voltage V out  from the charge pump output stage  26  and controls the frequency of the oscillator  24  utilizing the ratio of the input voltage V in  to the output voltage V out . 
     FIG. 3 is a block diagram of a preferred embodiment frequency control circuit  22  of the present invention. The frequency control circuit  22  comprises a first voltage-to-current converter  32  and a second voltage-to-current converter  34 . An input voltage V in  and output voltage V out  of the charge pump output stage  26  are input into the first and second voltage-to-current converters  32  and  34 . The voltage-to-current converters  32  and  34  are matched, i.e. they are identical in structure and function in order to provide accurate measurement of the ratio between the input voltage V in  and the output voltage V out . 
     The input voltage V in  is input into the first voltage-to-current converter  32  and the output voltage V out  is input into the second voltage-to-current converter  34 . The input and output voltages, V in  and V out , are converted to first and second currents I (+)  and I (−) , respectively, which are proportional in magnitude to their corresponding voltages. The first voltage-to-current converter  32 , however, is connected to an inverter  36  that converts the first current I (+)  to an inverted current I INV . The first current I (+)  is inverted because the converters  32  and  34  each produce sink currents that cannot be compared. The inverter  36 , then, facilitates the production of the inverted current I INV  (i.e. a source current) that can be compared to the second current I (−)  (i.e. a sink current). The inverter  36  and second voltage-to-current converter  34  are coupled to a regulation circuit  38  that is in turn coupled to the VFO  24  (shown in FIG.  2 ). 
     The regulation circuit  38  measures the ratio between the input and output voltages V in  and V out , respectively, and varies the frequency of the VFO  24  to facilitate a small internal resistance, low output voltage ripple at full load, and low quiescent current while allowing the use of small and inexpensive external capacitors. At light loads, the magnitude of the output voltage is at or close to the magnitude of the input voltage. Since the load current is small there is no problem to run with a low frequency even with small external capacitors. As shown by equation (3), the quiescent current I Q  would be minimal because of the low frequency. Also the voltage ripple is minimized, in view of equation (2), because the small load current offsets the low clock frequency and small output capacitance (produced by the small external capacitors). The resistance is not an issue because only a small percentage of voltage is lost since the load current is small and the product of load current and internal resistance is small as well. 
     However, when the load current increases, the output voltage tends to drop; thus the difference between the output voltage and input voltage increases as well. To compensate for this reduction in output voltage, the regulation circuit  38  signals the VFO  24  to run faster and thus run the output stage (i.e. converter) at an efficient level. At full load current the quiescent current is not an issue because it is just a small percentage of the load current and therefore generates only a small degradation of the efficiency. Thus the system runs with maximum frequency to guarantee a small internal resistance and a low output voltage ripple. The VFO  24  is coupled to an output stage that will provide the desired output as dictated by the application in which the device is used. 
     FIG. 4 is a schematic of a preferred embodiment of the present invention. The first voltage-to-current converter  32  comprises a first transistor MN 1  and a second transistor MN 2 . The first and second transistors MN 1 , MN 2  have gate nodes  40  that are coupled together and source nodes  42  that are tied to a ground node GND. The second transistor MN 2  is coupled to the inverter  36 , represented in this preferred embodiment as a current mirror comprising two PMOS transistors MP 1  and MP 2 . A first resistor R 1  is coupled between the first transistor MN 1  and an input node  44  that receives the input voltage V in . 
     The matched second voltage-to-current converter  34  comprises a third transistor MN 11  and a fourth transistor MN 12  having gate nodes  46  that are coupled together and source nodes  48  that are tied to the ground node GND. A second resistor R 11  is coupled between the third transistor and the output node  50  that receives the output voltage V out . The second resistor R 11  is preferably matched in value to the first resistor R 1 . If, for example, first resistor R 1  is smaller than second resistor R 11 , then second resistor R 11  will produce an additional offset current which should not be generated when the input voltage is equal to the output voltage (i.e. there is no load current). The additional offset current will operate as a load current and cause the frequency control circuit  22  to operate improperly. 
     The inverter  36  produces the inverted current I inv . The inverted current I INV  and second current I (−)  are summed to produce a net current I net . Because the inverted current I INV  and second current I (−)  are of a magnitude that is proportional to their respective voltages, the net current I net  is of a magnitude that represents the difference of the input and output voltages V in  and V out . 
     The net current is input into a preferred embodiment regulation circuit  38  that comprises multiple transistors, MN 3 -MN 7  and MP 3 . The net current I net  is formed on the drain of transistor MN 3 . The regulation circuit  38  further comprises current sources that represent a maximum current I max  and a minimum current I min  that will be provided to the VFO  24 . The minimum and maximum currents I min  and I max  provide the frequency sweep range of the VFO  24  that allows the converter to run efficiently with decreased loss of energy. The frequency of the VFO  24  is limited to a minimum frequency where the input and output voltages V in  and V out  are equal, that is the net current I net  is zero. If there was no designed minimum frequency, the VFO  24  would stop oscillating when the net current I net  was zero. Current source  52  prevents this by supplying the minimum current I min . The VFO  24  is also limited to a maximum frequency to prevent the VFO  24  from running at frequencies at which no benefits are gained in efficiency and energy is loss. 
     Transistor MN 5  is coupled to transistor MN 3  to form a current mirror. Transistor MN 5  operates to add the net current I net  to the minimum current I min . The resulting current is the control current I cont  that is provided to run the VFO  24  at a rate that is proportional in magnitude to the ratio of the input and output voltages V in  and V out . Transistor MN 3  also forms a current mirror with transistor MN 4 . If transistors MN 3  and MN 4  are the same size, then the current on the drain of transistor MN 4  is the same as that on transistor MN 3 . Thus, the net current I net  is mirrored onto the drain of transistor MN 4 . The drain of transistor MN 4  is coupled to the gate of transistor MP 3  allowing the net current I net  to be compared with the maximum current I max . If the net current I net  is bigger than the maximum current I max , then the gate of transistor MP 3  becomes negative with respect to its source, MP 3  switches on and turns on transistor MN 6  (i.e. forces current via transistor MN 6 ). Transistor MN 6  turns on transistor MN 7  that in turn sinks a portion of the net current I net  from transistor MN 3  such that the addition of the net current I net  and the minimum current I min  never exceeds the maximum current I max . In a preferred embodiment, the VFO  24  minimum frequency is 50 khz and maximum frequency is 500 khz. 
     Furthermore, the size of transistors MN 3  and MN 5  can be chosen to provide the necessary control loop sensitivity for the regulation circuit  38 . If the VFO  24  sensitivity is not compatible with the sensitivity of the regulation circuit  38  (i.e. the VFO  24  can not detect the changes in control current produced by the regulation circuit  38 ), the net current can be amplified by a factor necessary to provide correct operation. For example, by choosing the size of transistor MN 5  to be 5 times the size of MN 3 , then the sensitivity of the regulation circuit is increased by a factor of 5. Thus, the bigger the size of MN 5 , the smaller the output voltage drop you need for the same frequency modulation. However, caution must be taken in matching the voltage-to-current converters  32  and  34  to ensure that the sensitivity of the regulation circuit  38  is not bigger than a matching error, otherwise the frequency control circuit  22  will not operate correctly. 
     Although the present invention and its advantages have been described in detail, it should be understood that various changes, substitutions and alterations can be made herein without departing from the spirit and scope of the invention as defined by the appended claims. Moreover, the scope of the present application is not intended to be limited to the particular embodiments of the process, manufacture, means, methods and steps described in the specification. As one of ordinary skill in the art will readily appreciate from the disclosure of the present invention, processes, machines, manufacture, compositions of matter, means, methods, or steps, presently existing or later to be developed, that perform substantially the same function or achieve substantially the same result as the corresponding embodiments described herein may be utilized according to the present invention. Accordingly, the appended claims are intended to include within their scope such processes, machines, manufacture, compositions of matter, means, methods, or steps.