Abstract:
A disk drive is disclosed comprising a voice coil motor (VCM) for actuating a head radially over a disk. A VCM driver comprises first and second transistors forming a common collector amplifier for sensing a current flowing through a voice coil of the VCM. The common collector amplifier improves the accuracy of the current sense measurement when the VCM is driven in a pulse width modulated (PWM) mode.

Description:
CROSS REFERENCE TO RELATED APPLICATIONS AND PATENTS 
     This application is related to co-pending U.S. Patent Application Ser. No. 10/376,821 entitled “DISK DRIVE COMPRISING OSCILLATORS AND COUNTERS FOR SENSING CURRENT IN A VOICE COIL MOTOR” filed on Feb. 28, 2003, the disclosure of which is incorporated herein by reference. 
    
    
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to disk drives for computer systems. More particularly, the present invention relates to a disk drive comprising current sense circuitry for a voice coil motor (VCM). 
     2. Description of the Prior Art 
       FIG. 1  shows a prior art disk drive comprising a disk  2  rotated about a center axis by a spindle motor (not shown). A head  4  attached to a distal end of an actuator arm  6  is actuated radially over the disk  2  by a voice coil motor (VCM)  8 . The VCM  8  comprises a voice coil  10  which interacts with permanent magnets of a VCM yoke in order to rotate the actuator arm  6  about a pivot. The VCM  8  is typically driven in either a linear mode or in a pulse width modulated (PWM) mode. In addition, the motion of the VCM  8  may be controlled using a current feedback loop by sensing the amount of current flowing through the voice coil  10  which is proportional to the amount of torque applied to the actuator arm  6 . 
       FIG. 1  also shows a VCM driver  12  comprising a conventional H-bridge driver for driving the voice coil  10  shown as a resistance Rvcm  14  and an inductance Lvcm  16 . The H-bridge driver comprises a plurality of driver switches  18 A- 18 D for selectively connecting the ends of the voice coil  10  to a supply voltage  20  or to ground  22  depending on the desired direction of rotation. A plurality of diodes D 1 -D 4  protect the driver switches  18 A- 18 D from flyback currents generated from driving an inductive load. 
     In order to control the motion of the VCM  8  accurately using a current feedback loop it is important to measure the total integral of the current flowing through the voice coil  10 . Several problems arise when attempting to use the conventional techniques for sensing the current flowing through the voice coil  10  when driven in a PWM mode. Referring again to  FIG. 1 , if a single sense resistor Rsense  24  in series with the voice coil  10  is used to sense current, the PWM voltage appears on both sides of the resistor Rsense  24  at several volts at very high slew rates. This chop voltage (a square wave) must be rejected by sense amplifier  31  so that the very small voltage across Rsense  24  can be accurately measured. This high frequency AC voltage capacitively couples into the sense amplifier  31 , and creates offsets and nonlinearities which distort the current sense measurement. This problem exacerbates as the frequency of the PWM increases. 
     Another prior art current sensing technique uses a sense resistor Rsensep  26  in series with the supply voltage  20  and an amplifier  28 , or a sense resistor Rsensem  30  in series with ground  22  and an amplifier  32 . This technique avoids the common mode voltage problem associated with sense resistor Rseries  24 , however, it also leads to other problems related to inductive flyback currents. Assume, for example, that current is flowing to the right through the voice coil  10 . Initially, driver switches  18 A and  18 D are on, allowing Vpwr  20  to source the current through the sense resistors Rsensep  26  or Rsensem  30 . Driver switch  18 A is driven by a PWM signal, while driver switch  18 D is left on continually. When the PWM signal turns driver switch  18 A off, the inductive load keeps current flowing to the right in the coil regardless of the voltage applied because of the magnetic flux stored in the coil and its magnetic structure. This inductive current can cause diode D 2  or driver switch  18 B to conduct current, depending on the ratio of impedances. Since current is also flowing through switch  18 D, the flyback current momentarily cancels the current through sense resistor Rsensem  30 , which can distort the current sense measurement by creating a blank spot in the voltage waveform. Additionally, if the two halves of the H-bridge are switched alternately, flyback current from the inductive current can drive the voltage at the top of sense resistor Rsensem  30  below ground. When this happens, substrate parasitic transistors (shown as parasitic transistor  31  in  FIG. 1 ) are activated, drawing current from elsewhere in the driver circuitry in a random manner, both distorting the current measurement with this additional current and disrupting operation of the driver circuitry. 
     Regardless of how the H-bridge PWM switching is timed, shootthrough currents (caused by a brief simultaneous conduction between driver switches  18 A and  18 B or driver switches  18 C and  18 D) or gate charge injections can also create false values for current that distort the true measurement. These problems are generally avoided using sample/hold circuits  34  and  36 , which sample the voltage across the resistors  26  and  30  at a point in the PWM chop cycle where distortions due to flyback, shootthrough, switching, or diode conduction, do not occur. However, the sampling process adds delay to the loop. Additionally, an abrupt change from a large current to a small current leaves a time related sample distortion in the waveform that can be larger than the actual voltage value corresponding to the small current. The control system spends time responding to these spurious distortions which cause unwanted motion in the VCM. Still further, the sense amplifiers  28  and  32  must be designed such that their inputs can be driven below ground, or above the power supply, respectively, in order to sense current of all polarities. Sensing current above or below the power supply rails significantly increases the circuit complexity of a monolithic IC sense amplifier due to substrate current injection, which also removes current from the sense resistor in a nonlinear manner. 
     There is, therefore, a need to accurately sense the current flowing through the voice coil of disk drive VCM in order to implement a current feedback loop while driving the VCM in a PWM mode. 
     SUMMARY OF THE INVENTION 
     The present invention may be regarded as a disk drive comprising a disk, an actuator arm, a head connected to a distal end of the actuator arm, a voice coil motor (VCM) comprising a voice coil, the VCM for rotating the actuator arm about a pivot to actuate the head radially over the disk, and a VCM driver. The VCM driver comprises an H-bridge driver comprising a plurality of driver switches for driving current from a supply voltage through the voice coil to ground. A first sense resistor is connected in series between the supply voltage and at least one of the driver switches, and a second sense resistor is connected in series between at least one of the driver switches and ground. The VCM driver further comprises a first transistor having a first transistor terminal, a second transistor terminal, and a gate terminal, and a second transistor having a first transistor terminal, a second transistor terminal, and a gate terminal. A third sense resistor has a first end connected to a node between the first sense resistor and the at least one of the driver switches and a second end connected to the first transistor terminal of the first transistor. A fourth sense resistor has a first end connected to a node between the second sense resistor and the at least one of the driver switches and a second end connected to the first transistor terminal of the second transistor. A first gain resistor has a first end connected to at least the second transistor terminal of the first transistor, and a second end connected to a reference voltage. When the gate terminals of the first and second transistors are forward biased, a voltage across the gain resistor represents the current flowing through the voice coil. 
     In one embodiment, the VCM driver further comprises a current source for generating a current flowing through the first transistor, for example by connecting a resistor between the supply voltage and the first transistor terminal of the first transistor. In another embodiment, the VCM driver further comprises a current source for generating a current flowing through the second transistor, for example by connecting a resistor between the first transistor terminal of the second transistor and ground. 
     In one embodiment, the first and second transistors are bipolar junction transistors, and in an alternative embodiment, the first and second transistors are field effect transistors. 
     In yet another embodiment, the VCM driver further comprises a second gain resistor having a first end connected to the second transistor terminal of the second transistor, and a second end connected to the reference voltage. When the gate terminals of the first and second transistors are forward biased a voltage across the second gain resistor represents the current flowing through the voice coil. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  shows prior art techniques for sensing the current flowing through the voice coil of a disk drive VCM. 
         FIG. 2A  shows a disk drive according to an embodiment of the present invention comprising a VCM driver employing first and second transistors forming a common collector amplifier for sensing the current flowing through the voice coil of the VCM. 
         FIG. 2B  shows an embodiment for generating the bias voltages for the first and second transistors of FIG.  2 A. 
         FIG. 3A  shows a disk drive according to an embodiment of the present invention wherein the VCM driver further comprises first and second current sources for optimizing the operating mode of the first and second transistors. 
         FIG. 3B  shows an embodiment of the present invention wherein the first and second current sources of  FIG. 3A  are implemented using resistors. 
         FIG. 4  shows a disk drive according to an embodiment of the present invention wherein the VCM driver further comprises auto-zero circuitry to compensate for drift. 
         FIG. 5  shows voltage waveforms representing the current sense signals relative to the PWM signal controlling the driver switches. 
         FIG. 6  shows a disk drive according to an embodiment of the present invention wherein the VCM driver comprises auto-zero circuitry and a first and second current source for optimizing the operating mode of the first and second transistors. 
     
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       FIG. 2A  shows a disk drive according to an embodiment of the present invention comprising a disk  40 , an actuator arm  42 , a head  44  connected to a distal end of the actuator arm  42 , a voice coil motor (VCM)  46  comprising a voice coil  48 , the VCM  46  for rotating the actuator arm  42  about a pivot to actuate the head  44  radially over the disk  40 . A VCM driver  50  comprises an H-bridge driver having a plurality of driver switches  52 A- 52 D for driving current from a supply voltage  54  through the voice coil  48  to ground  56 . A first sense resistor Rsensep  58  is connected in series between the supply voltage  54  and at least one of the driver switches  52 A- 52 D, and a second sense resistor Rsensem  60  is connected in series between at least one of the driver switches  52 A- 52 D and ground  56 . The VCM driver  50  further comprises a first transistor  62  having a first transistor terminal, a second transistor terminal, and a gate terminal, and a second transistor  64  having a first transistor terminal, a second transistor terminal, and a gate terminal. A third sense resistor Rsp  66  has a first end connected to a node between the first sense resistor Rsensep  58  and the at least one of the driver switches  52 A- 52 D and a second end connected to the first transistor terminal of the first transistor  62 . A fourth sense resistor Rsm  68  has a first end connected to a node between the second sense resistor Rsensem  60  and the at least one of the driver switches  52 A- 52 D and a second end connected to the first transistor terminal of the second transistor  64 . A gain resistor Rg  70  has a first end connected to at least the second transistor terminal of the first transistor  62 , and a second end connected to a reference voltage X*Vref  72 . When the gate terminals of the first and second transistors  62  and  64  are forward biased (by bias voltage Vbias  74  and  76 ), a voltage across the gain resistor Rg  70  represents the current flowing through the voice coil  48 . 
     The bias voltage Vbias  74  and  76  sets the amount of bias current flowing through sense resistor Rsp  66  and sense resistor Rsm  68 . Since the currents through sense resistors Rsp  66  and Rsm  68  are substantially matched through the biasing arrangement, this bias current generates insignificant voltage on Rg  70 . 
     The supply voltage  54  may be supplied by any suitable source, such as from a host computer or generated internally during a power failure using the back EMF voltage of the spindle motor (not shown). 
     When driving the VCM  46  in a particular direction (e.g., moving the head  44  from the inner diameter toward the outer diameter of disk  40 ), a PWM signal (not shown) turns on driver switches  52 A and  52 D. Current flowing from the supply voltage  54  through the voice coil  48  to ground  56  generates a voltage drop across sense resistor Rsensep  58  which reduces the voltage across sense resistor Rsp  66  and therefore reduces the amount of current flowing through transistor  62 . Since the current flowing through transistor  64  does not change, the gain resistor Rg  70  sources the difference in current from the reference voltage X*Vref  72  and the voltage developed across the gain resistor Rg  70  represents the current flowing through the voice coil  48 . When the PWM signal turns off driver switches  52 A and  52 D and turns on driver switch  52 B and  52 C, current is forced by the inductance to flow from ground, through sense resistor Rsensem  60 , through driver switch  52 B, through the voice coil  48 , through driver switch  52 C to the supply voltage  54 . This flow generates a voltage drop below ground across sense resistor Rsensem  60  which increases the voltage across sense resistor Rsm  68  and therefore increases the amount of current flowing through transistor  64 . Since the current flowing through transistor  62  does not change, the gain resistor Rg  70  sources the difference in current from the reference voltage X*Vref  72  and the voltage developed across the gain resistor Rg  70  again represents the current flowing through the voice coil  48 . 
     When driving the VCM  46  in the opposite direction (e.g., moving the head  44  from the outer diameter toward the inner diameter of disk  40 ), the PWM signal turns on driver switches  52 C and  52 B. Current flowing from the supply voltage  54  through the voice coil  48  to ground  56  generates a voltage across sense resistor Rsensem  60  which reduces the voltage across sense resistor Rsm  68  and therefore reduces the amount of current flowing through transistor  64 . Since the current flowing through transistor  62  does not change, the gain resistor Rg  70  sinks the difference in current and the voltage developed across the gain resistor Rg  70  represents the current flowing through the voice coil  48 . When the PWM signal turns off driver switches  52 C and  52 B and turns on driver switch  52 A and  52 D, current is forced by the inductance to flow from ground, through driver switch  52 D, through the voice coil  48 , through driver switch  52 A, through sense resistor Rsensep  58  to the supply voltage  54 . This flow generates a voltage rise above the supply voltage  54  across sense resistor Rsensep  58  which increases the voltage across sense resistor Rsp  66  and therefore increases the amount of current flowing through transistor  62 . Since the current flowing through transistor  64  does not change, the gain resistor Rg  70  sinks the difference in current and the voltage developed across the gain resistor Rg  70  again represents the current flowing through the voice coil  48 . 
     If driver switches  52 A and  52 B momentarily cross conduct, a current spike flows through sense resistor Rsensep  58  and sense resistor Rsensem  60 . Since the current is identical in both sense resistors, and the resulting voltage spike generated across sense resistor Rsensep  58  and sense resistor Rsensem  60  is opposite in sign, sense resistor Rsp  66  and sense resistor Rsm  68  cause an identical change in current through transistors  62  and  64 . The result is a net zero change in the voltage across the gain resistor Rg  70 , and consequently the circuit rejects common mode currents, bias currents, and any current flowing through both sense resistors identically. The extent that the circuitry rejects common mode depends on the match between the various parts of the circuitry. This embodiment therefore generates an accurate voltage representation of the current regardless of the state of the driver switches  52 A- 52 D or the direction the current is flowing through the voice coil  48 , and suppresses secondary currents generated in the circuitry that do not actually flow through the voice coil  48 . 
       FIG. 2B  shows an embodiment of the present invention for generating the bias voltage Vbias  74  and  76  ( FIG. 2A ) applied to the gates of transistors  62  and  64 . A reference voltage vRef  78  is applied as the bias voltage  76  for transistor  64 . The reference voltage vRef  78  is also applied to transistor  80  to generate the bias voltage across resistors  82  and  84 . The voltage across resistor  84  is the bias voltage Vbias  74  for transistor  62 . A field effect transistor (MOSFET)  86  is used to buffer the voltage across the gain resistor Rg  70  representing the current flowing through the voice coil  48 . This buffer arrangement is very simple, fast, and very high impedance at its input. 
       FIG. 3A  shows an embodiment of the present invention wherein the VCM driver  50  further comprises a first current source  88  for optimizing the operating mode of the first transistor  62 , and a second current source  90  for optimizing the operating mode of the second transistor  64 . In this embodiment, the current sources  88  and  90  are configured so that the transistors  62  and  64  operate in a more linear region which improves the accuracy and speed of the current sense measurement (the voltage across the gain resistor Rg  70 ). The first and second current sources  88  and  90  may be implemented using any suitable circuitry, such as conventional current mirror circuits, or as shown in  FIG. 3B , by adding resistors  92  and  94 . Resistors  92  and  94  can be somewhat mismatched, adding an offset to the voltage across Rg  70 , but the AC portion of the current sense signal remains intact. 
       FIG. 4  shows an embodiment of the present invention wherein the VCM driver  50  comprises auto-zero circuitry to compensate for drift in the operating characteristics of the first and second transistors  62  and  64  (due, for example, to temperature drift or component mismatches). This embodiment employs first and second gain resistor Rgp  96  and Rgm  98  for generating a voltage with respect to X*Vref  72  representing the current flowing through the voice coil  48 . Transistors  100  and  102  have been added to provide bias currents to transistors  64  and  62 , respectively. Resistor Rb is approximately equal to sense resistors Rsp  66  and Rsm  68 , nominally zeroing the voltage across gain resistors Rgp  96  and Rgm  98  when zero current flows in the H-bridge. 
     When driving the VCM  46  in a particular direction (e.g., moving the head  44  from the inner diameter toward the outer diameter of disk  40 ), the PWM signal turns on driver switches  52 A and  52 D. Current flowing from the supply voltage  54  through the voice coil  48  to ground  56  generates a voltage drop across sense resistor Rsensep  58  which reduces the voltage across sense resistor Rsp  66  and therefore reduces the amount of current flowing through transistor  62 . Since the current flowing through transistor  102  does not change, the gain resistor Rgp  96  sources the difference in current from the reference voltage X*Vref  72  and the voltage developed across the gain resistor Rgp  96  represents the current flowing through the voice coil  48 . When the PWM signal turns off driver switches  52 A and  52 D and turns on driver switch  52 B and  52 C, current is forced by the inductance to flow from ground, through sense resistor Rsensem  60 , through driver switch  52 B, through the voice coil  48 , through driver switch  52 C to the supply voltage  54 . This flow generates a voltage drop below ground across sense resistor Rsensem  60  which increases the voltage across sense resistor Rsm  68  and therefore increases the amount of current flowing through transistor  64 . Since the current flowing through transistor  100  does not change, the gain resistor Rgm  98  sources the difference in current from the reference voltage X*Vref  72  and the voltage developed across the gain resistor Rgm  98  represents the current flowing through the voice coil  48 . 
     When driving the VCM  46  in the opposite direction (e.g., moving the head  44  from the outer diameter toward the inner diameter of disk  40 ), the PWM signal turns on driver switches  52 C and  52 B. Current flowing from the supply voltage  54  through the voice coil  48  to ground  56  generates a voltage across sense resistor Rsensem  60  which reduces the voltage across sense resistor Rsm  68  and therefore reduces the amount of current flowing through transistor  64 . Since the current flowing through transistor  100  does not change, the gain resistor Rgm  98  sinks the difference in current and the voltage developed across the gain resistor Rgm  98  represents the current flowing through the voice coil  48 . When the PWM signal turns off driver switches  52 C and  52 B and turns on driver switch  52 A and  52 D, current is forced by the inductance to flow from ground, through driver switch  52 D, through the voice coil  48 , through driver switch  52 A, through sense resistor Rsensep  58  to the supply voltage  54 . This flow generates a voltage rise above the supply voltage  54  across sense resistor Rsensep  58  which increases the voltage across sense resistor Rsp  66  and therefore increases the amount of current flowing through transistor  62 . Since the current flowing through transistor  102  does not change, the gain resistor Rgp  96  sinks the difference in current and the voltage developed across the gain resistor Rgp  96  represents the current flowing through the voice coil  48 . 
     A first and second MOSFETs  104  and  106  are used to buffer the respective voltages across the gain resistors Rgp  96  and Rgm  98  representing the current flowing through the voice coil  48 . 
     Since the driver switches  52 A- 52 D in the H-bridge driver are driven with a PWM signal, there is a known period of time during the PWM cycle when zero current is flowing through sense resistor Rsensep  58  and a known period of time when zero current is flowing through sense resistor Rsensem  60 . The voltage across the gain resistors Rgp  96  and Rgm  98  during these time intervals, which represents zero current, is used to adjust the voltage measurements when current is flowing through the gain resistors Rgp  96  and Rgm  98 . This auto-zero cycle compensates for drift in the operating characteristics of the entire sense circuit. It also allows the use of crude depletion mode MOSFETs  104  and  106  that need not be matched since the auto-zero cycle calibrates out the voltage difference. 
       FIG. 5  shows the voltage waveforms  108  and  110  across gain resistors Rgp  96  and Rgm  98  relative the PWM signal  112  controlling the driver switches  52 A- 52 D. The waveforms illustrate that the operating characteristics of transistors  62  and  64  may drift creating an offset in the voltage measurement across the gain resistors Rgp  96  and Rgm  98 . In this example when the PWM signal  112  is high, driver switches  52 B and  52 C are turned on and driver switches  52 A and  52 D are turned off. Zero current flows through sense resistor Rsensep  58  such that the voltage  108  across gain resistor Rgp  96  represents the zero-level offset voltage, while the voltage  110  across gain resistor Rgm  98  represents the zero-level offset voltage plus the current flowing through the voice coil  48 . When the PWM signal  112  is low, driver switches  52 A and  52 D are turned on and driver switches  52 B and  52 C are turned off. Zero current flows through sense resistor Rsensem  60  such that the voltage  110  across gain resistor Rgm  98  represents the zero-level offset voltage, while the voltage  108  across gain resistor Rgp  96  represents the zero-level offset voltage plus the current flowing through the voice coil  48 . The voltage  108  across gain resistor Rgp  96  while the PWM signal  112  is high is subtracted from the voltage  108  across gain resistor Rgp  96  while the PWM signal  112  is low. Similarly, the voltage  110  across the gain resistor Rgm  98  while the PWM signal  112  is low is subtracted from the voltage  110  across the gain resistor Rgm  98  while the PWM signal  112  is high. The resulting composite signal  114  represents the current following through the voice coil  48  with the offset voltage canceled. Any suitable circuitry may be employed to subtract the offset from voltages  108  and  110 , including sample/hold circuitry operating relative to the PWM cycle. In an alternative embodiment, oscillators and counters are used to subtract the offset from voltages  108  and  110  as disclosed in the above-referenced U.S. patent application entitled “DISK DRIVE COMPRISING OSCILLATORS AND COUNTERS FOR SENSING CURRENT IN A VOICE COIL MOTOR”. 
       FIG. 6  shows an embodiment wherein separate bias current adjustments are included by adding resistors  116  and  118  in order to optimize the operating mode (improve linearity and speed) of the first and second transistors  62  and  64  by increasing the idle current and thus reducing the transistor&#39;s internal impedances. 
     Any suitable transistor technology may be employed to implement transistors  62 ,  64 ,  100 ,  85 ,  102  and  80 . In the embodiments described above, the first and second transistors are bipolar junction transistors (BJT) wherein the first transistor  62  is a pnp BJT and the second transistor  64  is a npn BJT. In an alternative embodiment, the first and second transistors  62  and  64  comprise field effect transistors (FETs). Since the first and second transistors  62  and  64  are emitter (or source) driven, the current sense circuitry operates extremely fast (high bandwidth) with high fidelity.