Abstract:
In a precharge operation, when an external supply potential is lower than the lower limit determined by a specification, a VDC circuit in a sense amplifier operating voltage generating circuit supplies to a sense power supply line a potential equal to the external supply potential. When the external supply potential is higher than the lower limit determined by the specification, the VDC circuit supplies a potential equal to the lower limit of the external supply potential. Accordingly, a semiconductor integrated circuit device including this circuitry can achieve power savings without decrease in the sensing operation rate and without supply of charges more than necessary to memory cells.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to semiconductor integrated circuit devices. More specifically, the invention relates to a semiconductor integrated circuit device including a circuit for generating an internal supply voltage to be provided to a sense amplifier. 
     2. Description of the Background Art 
     The operating supply voltage of recent semiconductor integrated circuit devices has remarkably been decreasing. As an example, an array operating potential Vdds is now considered that is an operating supply potential of a sense amplifier and is equal to H data written into a memory cell of a dynamic random access memory (DRAM). 
     In general, array operating potential Vdds is generated by internally decreasing an external supply potential ext.Vdd. Array operating potential Vdds is determined from the reliability of an insulating film which constitutes a memory cell capacitor. The recent reduction in the design rule leads to reduction in the thickness of the insulating film. Then, decrease of a potential difference applied to the film is required. Accordingly, there arises a need for decrease of array operating potential Vdds because of the reduced thickness of the insulating film. 
     However, in terms of an array operating margin, the lowered level of array operating potential Vdds is disadvantageous. 
     FIG. 7 is a circuit diagram showing a partial structure of a memory cell array in a DRAM. 
     Referring to FIG. 7, the memory cell array in the DRAM includes a sense amplifier  30 , a bit line equalize circuit  20 , and a memory cell  10 . 
     Sense amplifier  30  includes P channel MOS transistors P 1  and P 2  and N channel MOS transistors N 1  and N 2 . 
     P channel MOS transistor P 1  is connected between a node A 3  and a P channel MOS transistor P 3 , and P channel MOS transistor P 2  is connected between a node A 4  and P channel MOS transistor P 3 . 
     N channel MOS transistor N 1  is connected between node A 3  and an N channel MOS transistor N 3 , and N channel MOS transistor N 2  is connected between node A 4  and N channel MOS transistor N 3 . 
     P channel MOS transistor P 1  and N channel MOS transistor N 1  have respective gates connected to node A 4  and P channel MOS transistor P 2  and N channel MOS transistor N 2  have respective gates connected to node A 3 . Node A 3  is connected to a bit line BL and node A 4  is connected to a bit line ZBL. 
     The source of P channel MOS transistor P 3  is connected to an internal supply voltage generating circuit (VDC)  40  via a sense power supply line VSH (interconnect resistance R 1 ) and the gate thereof is connected to a node ZSOP. 
     N channel MOS transistor N 3  is grounded via a node VSL (interconnect resistance R 2 ). 
     Bit line equalize circuit  20  includes an N channel MOS transistor N 4  connected between bit lines BL and ZBL and N channel MOS transistors N 5  and N 6  connected in series between bit lines BL and ZBL. Respective gates of N channel MOS transistors N 4  to N 6  are connected to a node A 2 . The connecting point of N channel MOS transistors N 5  and N 6  is connected to a node A 1 . Node A 2  receives a bit line equalize signal BLEQ and node A 1  receives a bit line potential Vbl. Bit line equalize circuit  20  equalizes the potentials on bit lines BL and ZBL to bit line potential Vbl in response to rising of bit line equalize signal BLEQ to H level of an activation level. Bit line potential Vbl is equal to half of array operating potential, Vdds/2. 
     Memory cell  10  includes an N channel MOS transistor N 7  for access and a capacitor C 1  for information storage. The gate of N channel MOS transistor N 7  in memory cell  10  is connected to a word line WL of a corresponding row. N channel MOS transistor N 7  is connected between bit line BL and one electrode (storage node SN) of capacitor C 1 . The other electrode of capacitor C 1  receives a cell plate potential Vcp. Word line WL activates memory cell  10 . Paired bit lines BL and ZBL supply/receive a data signal to and from a selected memory cell. 
     When memory cell  10  holds H data, a data reading operation is performed as described below. 
     FIG. 8 is a timing chart showing an operation of sense amplifier  30  in FIG.  7 . 
     Referring to FIG. 8, in a precharge state prior to time T 1 , bit line equalize signal BLEQ in bit line equalize circuit  20  has H level and accordingly N channel MOS transistors N 4  to N 6  in bit line equalize circuit  20  are turned on. Then, before time T 1 , respective potentials on paired bit lines BL and ZBL are precharged to bit line potential Vbl which is the intermediate potential between array operating potential Vdds of an H data potential and ground potential GND of an L data potential. 
     At time T 1 , word line WL is activated to H level so that N channel MOS transistor N 7  in memory cell  10  is turned on and the H data held in memory cell  10  is transmitted to bit line BL. Consequently, the potential on bit line BL increases from bit line potential Vbl by a minute potential dV. The potential on bit line ZBL stays at bit line potential Vbl and thus a potential difference occurs between paired bit lines BL and ZBL. 
     At time T 2 , sense amplifier activation signals ZS 0 P and S 0 N become respectively to L and H levels so that P channel MOS transistor P 3  and N channel MOS transistor N 3  are turned on and sense amplifier  30  is activated. Then, the potential difference between paired bit lines BL and ZBL is amplified and bit line BL and storage node SN of memory cell  10  are raised to array operating potential Vdds which is the potential of H data. Moreover, the potential on bit line ZBL is lowered from bit line potential Vbl to ground potential GND. 
     It is supposed here that P channel MOS transistors P 1  and P 2  constituting sense amplifier  30  both have a threshold potential Vthp and N channel MOS transistors N 1  and N 2  constituting sense amplifier  30  both have a threshold potential Vthn. In order for sense amplifier  30  to start its operation at time T 2 , it is necessary that gate-source potential Vgs of P channel MOS transistors P 1  and P 2  should be higher than potential Vthp and gate-source potential Vgs of N channel MOS transistors N 1  and N 2  should be higher than potential Vthp. Gate-source potential Vgs can be represented by the following equation when minute potential dV is ignored. 
     
       
           Vgs=Vbl=Vdds /2 
       
     
     Then, for operation of sense amplifier  30 , array operating potential Vdds should have the relation below. 
     
       
           Vdds &gt;max(2 ×Vthn , 2 ×|Vthp |)  (1) 
       
     
     Accordingly, array operating potential Vdds on sense power supply line VSH should be any potential which satisfies relation (1). 
     Further, an initial operating speed of sense amplifier  30  is determined by respective differences, Vgs−|Vthp| and Vgs−Vthn, between gate-source potential Vgs of respective MOS transistors in sense amplifier  30  and threshold voltages Vthp and Vthn of respective MOS transistors. 
     In view of this, if threshold voltages Vthp and Vthn of respective transistors vary due to change in a manufacturing process, a decreased array operating potential Vdds results in an insufficient operation margin of sense amplifier  30 . In addition, if the decreased array operating potential Vdds makes it impossible to obtain an enough Vgs−|Vthp| or Vgs−Vthn, sense amplifier  30  requires an extended operating time. 
     After time T 2  in FIG. 8, the potentials on sense power supply line VSH and node VSL during operation of sense amplifier  30  change transitionally depending on the interconnect resistances of sense power supply line VSH and node VSL, response rate of VDC circuit  40  and the like. In other words, the potential on sense power supply line VSH decreases to the lowest level at time T 3  and the potential on node VSL increases to the highest level at time T 3 . Such a variation of the potentials on sense power supply line VSH and node VSL during the sensing operation considerably deteriorates the operating speed of sense amplifier  30 . 
     In order to resolve the problem of insufficient operation margin of sense amplifier  30  due to the reduction of array operating potential Vdds, “overdrive method” is proposed as a method of supplying charges to sense power supply line VSH. 
     First Overdrive Method 
     One example of the overdrive method is described below that is proposed in Japanese Patent Laying-Open No. 11-250665 and Takasi Kono, 1999 Symposium on VLSI Circuits, Digest of Technical Papers, pp. 123-124. 
     FIG. 9 is a circuit diagram showing a partial structure of a memory cell array in a DRAM including a sense amplifier drive circuit according to the overdrive method. 
     Referring to FIG. 9, a sense amplifier operating voltage generating circuit  90  is provided instead of VDC circuit  40  in the circuit diagram of FIG.  7 . 
     FIG. 10 is a circuit diagram of sense amplifier operating voltage generating circuit  90  in FIG.  9 . 
     Referring to FIG. 10, sense amplifier operating voltage generating circuit  90  includes a reference potential generating circuit  100 , a selector circuit  150 , a shifter circuit  160 , a VDC circuit  170 , a P channel driver circuit  200  and a decoupling capacitor C 2 . 
     Reference potential generating circuit  100  includes a low-pass filter (LPF)  110  for eliminating noise on an external supply potential ext.Vdd, a constant current circuit  120 , and an output circuit  130  for outputting a predetermined voltage. Output circuit  130  includes a first reference potential output stage  131  and a second reference potential output stage  136 . 
     Low-pass filter  110  includes a resistor R 20  and a capacitor C 20  connected in series between an external supply node ext.Vdd and a ground node GND, and outputs to constant current circuit  120  a potential with noise on external supply potential ext.Vdd removed therefrom. 
     Constant current circuit  120  includes a P channel MOS transistor P 10  having its source connected to a node A 5  and its gate and drain connected to a node A 6 , an N channel MOS transistor N 10  connected between node A 6  and ground node GND and having its gate connected to a node A 7 , a resistor R 21  connected between node A 5  and the source of a P channel MOS transistor P 11 , P channel MOS transistor P 11  connected between resistor R 21  and node A 7  and having its gate connected to node A 6 , and an N channel MOS transistor N 11  having its source connected to ground node GND and drain and gate connected to node A 7 . 
     Constant current circuit  120  generates a constant current Ir which does not depend on external supply potential ext.Vdd. 
     The first reference potential output stage  131  in output circuit  130  is constituted of P channel MOS transistors P 12  to P 15 . P channel MOS transistor P 12  simply supplies constant current Ir while P channel MOS transistors P 13  to P 15  operate as resistors. Then, the first reference potential output stage  131  outputs a potential Vrefs equal to an array operating potential Vdds. The second reference potential output stage  136  constituted of P channel MOS transistors P 16  to P 19  outputs a potential Vrefp equal to an internal potential Vddp used by peripheral circuitry of the memory cell array portion. 
     Selector circuit  150  includes a transfer gate  151  connected to the second reference potential output stage  136 , a transfer gate  152  connected to the first reference potential output stage  131 , and an inverter  153 . Transfer gates  151  and  152  have respective gates receiving a signal PRE for inactivating a row-related circuit to output potential Vrefp when signal PRE has H level and output potential Vrefs when signal PRE has L level. 
     Shifter circuit  160  includes an N channel MOS transistor N 20  connected between nodes A 10  and A 12  and having its gate receiving an output signal from selector circuit  150 , an N channel MOS transistor N 22  connected between node A 10  and ground node GND and having its gate connected to a node A 11 , an N channel MOS transistor N 21  connected between nodes A 12  and A 11  and having its gate receiving a potential on a sense power supply line VSH, and an N channel MOS transistor N 23  having its gate and drain connected to node A 11  and its source connected to ground node GND. A signal REF is output from node A 10  of shifter circuit  160  while a signal SIG is output from node A 11 . 
     VDC circuit  170  includes a comparator  180  constituted of a differential amplifier circuit and a P channel driver circuit  190  including a P channel MOS transistor P 22  connected to sense power supply line VSH and external supply node ext.Vdd. 
     Comparator  180  includes a P channel MOS transistor P 20  having its source connected to a node A 13  supplied with external supply potential ext.Vdd and having its gate and drain connected to a node A 14 , an N channel MOS transistor N 24  connected between nodes A 14  and A 16  and having its gate receiving signal SIG, a P channel MOS transistor P 21  connected between nodes A 13  and A 15  and having its gate connected to node A 14 , an N channel MOS transistor N 25  connected between nodes A 15  and A 16  and having its gate receiving signal REF, and an N channel MOS transistor N 26  connected between node A 16  and ground node GND and having its gate receiving external supply potential ext.Vdd. 
     P channel MOS transistor P 22  in P channel driver circuit  190  receives, at its gate, an output potential from comparator  180  and supplies the potential to sense power supply line VSH. 
     Signals SIG and REF supplied from shifter circuit  160  change in respective ranges centering on respective levels which are respectively almost a half of the potential from selector circuit  150  and almost a half of the potential on sense power supply line VSH. Then, N channel MOS transistors N 24  and N 25  in comparator  180  receiving these signals can operate in a saturation region even if the potential on node A 16  is close to the ground potential. As a result, gate-source potential Vgs of P channel MOS transistor P 22  in P channel driver circuit  190  can be increased. In other words, even if the transistor size of P channel MOS transistor P 22  is relatively small, the VDC circuit having a satisfactory current supply capability can be implemented. 
     P channel driver circuit  200  includes a P channel MOS transistor P 23  connected between an internal potential node Vddp and sense power supply line VSH and an inverter  202  connected to the gate of P channel MOS transistor P 23 . 
     P channel MOS transistor P 23  in P channel driver circuit  200  is turned on when signal PRE has H level to supply internal potential Vddp to sense power supply line VSH. 
     Sense amplifier operating voltage generating circuit  90  having the above circuit structure according to the overdrive method operates as discussed below. 
     FIG. 11 is a timing chart showing an operation of sense amplifier operating voltage generating circuit  90  shown in FIG.  10 . 
     Referring to FIG. 11, in a precharge state prior to time T 4 , signal PRE has H level so that an output signal supplied from selector circuit  150  is potential Vrefp. Then, internal potential Vddp equal to potential Vrefp is supplied from VDC circuit  170  to sense power supply line VSH at the time of precharge. At the same time, P channel MOS transistor P 23  in P channel driver circuit  200  is turned on so that internal potential Vddp is supplied from P channel driver circuit  200  to sense power supply line VSH. 
     Accordingly, when signal PRE has H level before time T 4 , sense power supply line VSH and decoupling capacitor C 2  are always supplied with internal potential Vddp at the time of precharge. 
     Prior to time T 4  when word line WL is activated, signal PRE becomes L level. Then, P channel MOS transistor P 23  in P channel driver  200  is turned off and accordingly sense power supply line VSH and decoupling capacitor C 2  are separated from internal potential Vddp. The potential output from selector circuit  150  is potential Vrefs so that the potential supplied from VDC circuit  170  to sense power supply line VSH is array operating potential Vdds. 
     At time T 5 , activation signals S 0 N and ZS 0 P become H and L levels respectively to start the operation of sense amplifier  30 . Then, charges accumulated on decoupling capacitor C 2  flow onto sense power supply line VSH. As a result, the potential on sense power supply line VSH decreases lower than array operating potential Vdds to a decreased extent and accordingly a higher rate of the sense amplify operation is achieved. 
     The capacitance of decoupling capacitor C 2  can appropriately be set to make the potential on sense power supply line VSH after completion of sensing operation equal to array operating potential Vdds which is H data potential. However, the potential supplied from VDC circuit  170  to sense power supply line VSH at time T 4  is array operating potential Vdds, therefore, even if sense power supply line VSH has its potential equal to or lower than array operating potential Vdds due to insufficient charges accumulated on the decoupling capacitor in the sensing operation, VDC circuit  170  supplies charges corresponding to the shortage of charges. In this way, the potential on sense power supply line VSH is kept at array operating potential Vdds. 
     Sense amplifier operating voltage generating circuit  90  having the circuit structure shown in FIG. 10 can be used to increase the rate of sensing operation in the initial stage relative to the conventional sense amplifier and thus a sufficient sense margin can be secured even at a low array operating potential Vdds. 
     The reason for the above advantage is that, in the initial stage of sensing operation by sense amplifier operating voltage generating circuit  90 , gate-source potential Vgs of each MOS transistor in sense amplifier  30  increases from the conventional (Vdds/2) to (Vddp−Vdds/2) by (Vddp−Vdds). 
     The sensing operation by sense amplifier operating voltage generating circuit  90  having the circuit structure shown in FIG. 10 is effective when external supply potential ext.Vdd, internal potential Vddp and array operating potential Vdds have the following relation: 
     external supply potential ext.Vdd&gt;internal potential Vddp&gt;array operating potential Vdds. 
     Second Overdrive Method 
     Another example of the overdrive method is disclosed as an overdrive sensing method in Japanese Patent Laying-Open No. 11-250665 described below. 
     FIG. 12 is a circuit diagram of a sense amplifier drive circuit according to the second overdrive method. 
     Referring to FIG. 12, a sense amplifier operating voltage generating circuit  300  includes a reference potential generating circuit  301  outputting a potential Vrefs equal to an array operating potential Vdds, a VDC circuit  306 , a P channel driver circuit  307  and a decoupling capacitor C 3 . 
     Reference potential generating circuit  301  generates reference potential Vrefs equal to array operating potential Vdds and supplies reference potential Vrefs to VDC circuit  306 . 
     VDC circuit  306  includes a comparator  302  and a P channel driver circuit  303 . Comparator  302  is a differential amplifier circuit constituted of P channel MOS transistors P 20  and P 21  and N channel MOS transistors N 24 , N 25  and N 26 . The gate of N channel MOS transistor N 24  receives the potential on a sense power supply line VSH while the gate of N channel MOS transistor N 25  receives reference potential Vrefs. P channel driver circuit  303  includes a P channel MOS transistor P 22  connected between an external supply potential ext.Vdd and sense power supply line VSH. 
     P channel driver circuit  307  includes a P channel MOS transistor P 30  which is connected between external supply node ext.Vdd and sense power supply line VSH and has its gate supplied with a signal φ. 
     Sense amplifier operating voltage generating circuit  300  having the above circuit structure operates as described below. 
     FIG. 13 is a timing chart showing an operation of sense amplifier operating voltage generating circuit  90  shown in FIG.  12 . 
     Referring to FIG. 13, prior to time T 6 , signal φ is at L level so that P channel MOS transistor P 30  is turned on and sense power supply line VSH is precharged to external supply potential ext.Vdd. 
     At time T 6 , sense amplifier activation signals S 0 N and ZS 0 P become respectively to H and L levels to start the operation of sense amplifier  30 . Then, each MOS transistor in sense amplifier  30  has its gate-source potential Vgs higher than the conventional one. Signal φ stays at L level until time T 7  and sense power supply line VSH is provided with external supply potential ext.Vdd and accordingly the sense amplifier operation is increased in rate. On the other hand, if there is a shortage of charges required for sensing operation after time T 7 , charges are supplied from VDC circuit  306  and accordingly the potential on sense power supply line VSH is maintained at array operating potential Vdds. 
     When sense amplifier operating voltage generating circuit  90  or  300  according to the overdrive method as described above is employed to perform a sensing operation, the potential on sense power supply line VSH in sensing operation never exhibits such a remarkable decrease as that occurs at time T 3  in FIG.  8 . Consequently, the rate of sensing operation can be increased. 
     However, a problem arises when, in a semiconductor integrated circuit including the sense amplifier operating voltage generating circuit of the overdrive method, external supply potential ext.Vdd is decreased for the purpose of saving power. 
     It is supposed here that external supply potential ext.Vdd to be provided to a semiconductor integrated circuit device including a sense amplifier operating voltage generating circuit of the overdrive method is reduced for saving power, and consequently the relation, external supply potential ext.Vdd=internal potential Vddp is established. 
     In this case, in sense amplifier operating voltage generating circuits  90  and  300  of the overdrive method, respective decoupling capacitors C 2  and C 3  being precharged are both supplied with external supply potential ext.Vdd. Here, external supply potential ext.Vdd varies in an allowable range defined by a specification. Then, the amount of charges accumulated on decoupling capacitors C 2  and C 3  being precharged varies. 
     If the amount of accumulated charges is smaller than that necessary for a sensing operation, an amount of charges corresponding to the shortage is provided from VDC circuits  170  and  306  and thus no problem occurs. However, if the variation of external supply potential ext.Vdd causes the amount of accumulated charges to be larger than a necessary amount of charges, the potential on sense power supply line VSH in sensing operation becomes higher than H data potential which is not preferable in terms of reliability of memory cells. 
     Specifically, suppose that capacitance of decoupling capacitors C 2  and C 3  is Cd, total amount of negative charges on bit line BL or ZBL is Cba, precharge potential on sense power supply line VSH is Vpre and precharge level of a bit line is Vbl (=Vdds/2), and the following relation is satisfied. 
     
       
           Cd ×( Vpre−Vdds )= Cba×Vbl   (2) 
       
     
     In this case, if external supply potential ext.Vdd is higher than precharge potential Vpre, the potential on sense power supply line VSH in sensing operation is higher than H data potential which is not preferable in terms of reliability. 
     SUMMARY OF THE INVENTION 
     One object of the present invention is to provide a semiconductor integrated circuit device achieving power savings without decrease in the operating rate of a sense amplifier and without supply of charges more than necessary to a memory cell. 
     A semiconductor integrated circuit device according to the present invention includes paired bit lines, a memory cell connected to one of the paired bit lines, a sense amplifier for amplifying a potential difference between the paired bit lines generated by reading of data from the memory cell, and a sense amplifier operating voltage generating circuit for supplying a voltage accumulated in the memory cell to the sense amplifier in an active period of the sense amplifier. The sense amplifier operating voltage generating circuit includes an internal potential supply node connected to the sense amplifier, a first voltage supply circuit for outputting, when an external supply voltage is higher than a predetermined voltage, the predetermined voltage as an output voltage to the internal potential supply node and outputting, when the external supply voltage is lower than the predetermined voltage, a voltage equal to the external supply voltage as an output voltage to the internal potential supply node, and a decoupling capacitor connected to the internal potential supply node. 
     Preferably, the first voltage supply circuit is stopped from operating in the active period of the sense amplifier. 
     Still preferably, the first voltage supply circuit includes a reference voltage generating circuit for outputting a voltage lower than the external supply voltage as a reference voltage, a shift circuit for reducing the output voltage to output the reduced voltage, and a voltage downconverter circuit receiving the reference voltage and the reduced voltage to output the output voltage. 
     Still more preferably, the voltage downconverter circuit includes a comparator circuit receiving the reference voltage and the reduced voltage to output a result of comparison between the reference voltage and the reduced voltage, and a switching element connected to an external supply node receiving the external supply voltage and the internal potential supply node, and the switching element receives the result of comparison from the comparator circuit to control the output voltage of the internal potential supply node. 
     Accordingly, it is possible to avoid charges more than necessary from being supplied to bit lines in a sense amplifier operation. 
     Still more preferably, the shift circuit includes a plurality of resistor elements connected in series between the internal potential supply node and a ground node. 
     The output voltage can thus be reduced. 
     Still more preferably, the shift circuit includes a first transistor and a second transistor connected in series between the external supply node and the ground node, and the output voltage is input to a control electrode of the first transistor and the reference voltage is input to a control electrode of the second transistor. 
     Then, variations of the output voltage can readily be adjusted due to changes of manufacture process of the semiconductor integrated circuit device. 
     Still more preferably, the sense amplifier operating voltage generating circuit further includes a second voltage supply circuit for outputting the predetermined voltage to the internal potential supply node when the output voltage held in the decoupling capacitor by charging is lower than the predetermined voltage in an inactive period of the sense amplifier. 
     Still more preferably, the second voltage supply circuit supplies the predetermined voltage in the active period of the sense amplifier. 
     Accordingly, a shortage of charges can be prevented that are to be supplied to bit lines in a sense amplifier operation. 
     Still more preferably, the predetermined voltage is a lower limit of the external supply voltage determined by a specification. 
     Then, excessive supply of charges to bit lines due to variations of the external supply voltage can be prevented. 
     According to the present invention, the semiconductor integrated circuit device can be provided that achieves power savings without decrease in the operating rate of the sense amplifier. 
     The foregoing and other objects, features, aspects and advantages of the present invention will become more apparent from the following detailed description of the present invention when taken in conjunction with the accompanying drawings. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a circuit diagram of a sense amplifier operating voltage generating circuit according to an embodiment of the present invention. 
     FIG. 2 is a circuit diagram of a reference potential generating circuit shown in FIG.  1 . 
     FIG. 3 is a circuit diagram of a comparator shown in FIG.  1 . 
     FIG. 4 shows a relation between a precharge potential Vpre on a sense power supply line VSH and an external supply potential ext.Vdd at the time of precharge when the sense amplifier operating voltage generating circuit is used. 
     FIG. 5 is a circuit diagram of a sense amplifier operating voltage generating circuit according to a second embodiment of the present invention. 
     FIG. 6 is a circuit diagram of a sense amplifier operating voltage generating circuit according to a third embodiment of the present invention. 
     FIG. 7 is a circuit diagram showing a partial structure of a memory cell array in a DRAM. 
     FIG. 8 is a timing chart showing an operation of a sense amplifier shown in FIG.  7 . 
     FIG. 9 is a circuit diagram showing a partial structure of a memory cell array in a DRAM including a sense amplifier drive circuit according to an overdrive method. 
     FIG. 10 is a circuit diagram of a sense amplifier operating voltage generating circuit in FIG.  9 . 
     FIG. 11 is a timing chart showing an operation of the sense amplifier operating voltage generating circuit shown in FIG.  10 . 
     FIG. 12 is a circuit diagram of a sense amplifier drive circuit according to a second overdrive method. 
     FIG. 13 is a timing chart showing an operation of a sense amplifier operating voltage generating circuit shown in FIG.  12 . 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Embodiments of the present invention are hereinafter described in detail in conjunction with the drawings. The same or corresponding components in the drawings are denoted by the same reference character and description thereof is not repeated. 
     First Embodiment 
     FIG. 1 is a circuit diagram of a sense amplifier operating voltage generating circuit according to a first embodiment of the present invention. 
     Referring to FIG. 1, sense amplifier operating voltage generating circuit  400  includes a reference potential generating circuit  401 , a comparator  402 , a P channel driver circuit  403 , a level shift circuit  404  and a decoupling capacitor C 10 . 
     P channel driver circuit  403  includes a P channel MOS transistor P 40 . The source of P channel MOS transistor P 40  is connected to an external supply node ext.Vdd and the drain thereof is connected to a sense power supply line VSH. The gate of P channel MOS transistor P 40  receives an output signal from comparator  402 . Decoupling capacitor C 10  is connected between sense power supply line VSH and a ground node GND. 
     Level shift circuit  404  is constituted of resistor elements R 3  and R 4  connected in series between sense power supply line VSH and ground node GND and outputs a potential from a node A 40  connecting resistor elements R 3  and R 4 , the potential generated by dividing the potential on sense power supply line VSH. 
     FIG. 2 is a circuit diagram of reference potential generating circuit  401  shown in FIG.  1 . 
     Referring to FIG. 2, reference potential generating circuit  401  includes a low-pass filter  411 , a constant current circuit  412 , and an output circuit  413  controlled by constant current circuit  412  to output a reference potential Vref. 
     Low-pass filter  411  includes a resistor R 11  and a capacitor C 11  connected in series between external supply node ext.Vdd and ground node GND. 
     Low-pass filter  411  receives an external supply potential ext.Vdd to remove noise therefrom and supplies a resultant potential to a node A 41 . 
     Constant current circuit  412  includes a P channel MOS transistor P 41  having its source connected to node A 41  and gate and drain connected to a node A 42 , an N channel MOS transistor N 41  connected between node A 42  and ground node GND and having its gate connected to a node A 43 , a resistor R 12  connected between node A 41  and the source of a P channel MOS transistor P 42 , P channel MOS transistor P 42  connected between resistor R 12  and node A 43  and having its gate connected to node A 42 , and an N channel MOS transistor N 42  having its source connected to ground node GND and drain and gate connected to node A 43 . 
     N channel MOS transistors N 41  and N 42  form a current mirror circuit. Respective current driving capabilities of N channel MOS transistors N 41  and N 42  are made equal so that the same current I flows through two branches between external supply node ext.Vdd and ground node GND. Moreover, P channel MOS transistors P 41  and P 42  are operated in a subthreshold region and the current driving capability of P channel MOS transistor P 41  is made smaller than that of P channel MOS transistor P 42 . Then, current I is a constant current independent of external supply potential ext.Vdd. 
     Output circuit  413  includes a P channel MOS transistor P 43  connected between external supply node ext.Vdd and a node A 44  and having its gate connected to node A 42  of constant current circuit  412 , P channel MOS transistors P 44  and P 45  connected in series between nodes A 44  and A 45  and having respective gates both connected to node A 45 , and a P channel MOS transistor P 46  having its source connected to node A 45  and gate and drain connected to ground node GND. 
     P channel MOS transistor P 43  forms a current mirror together with constant current circuit  412  and accordingly current I is correctly duplicated. P channel MOS transistors P 44  and P 45  have channel resistance components and P channel MOS transistor P 46  has a threshold component which is a resistance component determined chiefly by a threshold. 
     Positive temperature characteristics of current I are reflected in the channel resistance components while the threshold component has negative temperature characteristics. Accordingly, these characteristics can be balanced to eliminate temperature characteristics of reference potential Vref. 
     FIG. 3 is a circuit diagram of comparator  402  shown in FIG.  1 . 
     Referring to FIG. 3, comparator  402  includes a P channel MOS transistor P 51  having its source connected to a node A 51  provided with external supply potential ext.Vdd and its gate and drain connected to a node A 52 , an N channel MOS transistor N 51  connected between nodes A 52  and A 54  and having its gate receiving a shift output potential SVpre supplied from level shift circuit  404 , a P channel MOS transistor P 52  connected between nodes A 51  and A 53  and having its gate connected to node A 52 , an N channel MOS transistor N 52  connected between nodes A 53  and A 54  and having its gate receiving reference potential Vref, and an N channel MOS transistor N 53  connected between node A 54  and ground node GND and having its gate receiving a signal PRE. 
     When signal PRE has H level, N channel MOS transistor N 53  is turned on and accordingly comparator  402  outputs a signal DO from node A 53 . When signal PRE has L level, comparator  402  stops its operation. 
     Sense amplifier operating voltage generating circuit  400  operates as described below when external supply potential ext.Vdd is reduced for power saving and consequently becomes equal to an internal potential Vddp. Here, an array operating potential Vdds equal to an H data potential written into a memory cell is a constant potential lower than a potential ext.Vdd (min) which is the lower limit of external supply potential ext.Vdd defined by a specification. 
     Reference potential Vref supplied from reference potential generating circuit  401  is set so that the relation represented by the following equation is established between reference potential Vref and potential ext.Vdd (min) which is the lower limit of external supply potential ext.Vdd defined by a specification. 
     
       
           Vref =α×ext. Vdd  (min)  (3) 
       
     
     Here, α is smaller than 1 (α&lt;1). Namely, reference potential Vref is made smaller than potential ext.Vdd (min). The reason is that, if reference potential Vref is made equal to potential ext.Vdd (min) in reference potential generating circuit  401  and external supply potential ext.Vdd is actually close to potential ext.Vdd (min), it is difficult for circuitry to generate a constant reference potential Vref which is independent of external supply potential ext.Vdd. In addition, there may be a case in which external supply potential ext.Vdd becomes smaller than the lower limit potential ext.Vdd (min) defined by a specification for some reason such as a transient decrease of external supply potential ext.Vdd in use. Then, reference potential generating circuit  401  receiving a potential from its power source, i.e., external supply potential ext.Vdd, cannot output a potential higher than external supply potential ext.Vdd. Accordingly, the relation, reference potential Vref&lt;potential ext.Vdd (min) is established. In order to adapt to a greater variation of external supply potential ext.Vdd, reference potential Vref is preferably set smaller than potential ext.Vdd (min). 
     The relation between a precharge potential Vpre and reference potential Vref is set as represented by the following equation, where precharge potential Vpre is a potential output to sense power supply line VSH from P channel driver circuit  403  in a precharge operation. 
     
       
           Vpre=Vref/α   (4) 
       
     
     Further, resistance values of resistors R 3  and R 4  in level shift circuit  404  are set so that shift output potential SVpre supplied from level shift circuit  404  satisfies the relation represented by the following equation. 
     
       
           SVpre=α×Vpre   (5) 
       
     
     A relation is now described between variation of external supply potential ext.Vdd and precharge potential Vpre supplied from VDC circuit  410  at the time of precharge. 
     At the time of precharge, signal PRE supplied to comparator  402  in VDC circuit  410  has H level. Comparator  402  then receives reference potential Vref determined by equation (3) and shift output potential SVpre determined by equation (5) to output to P channel driver circuit  403  signal DO according to the potential difference between reference potential Vref and shift output potential SVpre. In response to variation of external supply potential ext.Vdd, P channel driver circuit  403  in VDC circuit  410  outputs precharge potential Vpre to sense power supply line VSH as detailed below. 
     (1) External supply potential ext.Vdd is smaller than potential ext.Vdd (min) which is the lower limit of external supply potential ext.Vdd defined by a specification. 
     In this case, precharge potential Vpre supplied from P channel driver  403  is lower than potential ext.Vdd (min) and accordingly shift output potential SVpre output from level shift circuit  404  is always smaller than reference potential Vref. 
     As a result, the potential of signal DO supplied from comparator  402  decreases and P channel MOS transistor P 40  in P channel driver circuit  403  is constantly made on. 
     Accordingly, precharge potential Vpre output from P channel driver circuit  403  is always equal to external supply potential ext.Vdd. Then, external supply potential ext.Vdd is supplied to decoupling capacitor C 10  in a precharge operation. 
     (2) External supply potential ext.Vdd is higher than potential ext.Vdd (min). 
     In this case, if charging by P channel driver circuit  403  causes precharge potential Vpre to be higher than potential ext.Vdd (min), shift output potential SVpre output from level shift circuit  404  is higher than reference potential Vref. 
     The potential of signal DO supplied from comparator  402  accordingly increases so that P channel MOS transistor P 40  is turned off. 
     P channel driver circuit  403  is thus controlled to make precharge potential Vpre on sense power supply line VSH equal to potential ext.Vdd (min). 
     Then, precharge potential Vpre on sense power supply line VSH is always equal to potential ext.Vdd (min) and potential ext.Vdd (min) is supplied to decoupling capacitor C 10  in a precharge operation. 
     When sense amplifier operating voltage generating circuit  400  shown in FIG.  1  and operating as described above is used, precharge potential Vpre on sense power supply line VSH and external supply potential ext.Vdd have a relation as shown in FIG.  4 . 
     From the relation shown in FIG. 4 between precharge potential Vpre and external supply potential ext.Vdd and equation ( 2 ), it is possible to determine a capacitance Cd of decoupling capacitor C 10  in FIG. 1 by the following equation. 
     
       
           Cd=Cba×Vble /(ext. Vdd (min)− Vdds )  (6) 
       
     
     In a sensing operation, signal PRE supplied to the comparator is off and accordingly sense amplifier operating voltage generating circuit  400  shown in FIG. 1 is stopped from operating. Then, in the sensing operation, charges accumulated on decoupling capacitor C 10  during a precharge operation are supplied to sense power supply line VSH in order to prevent decrease in the operating rate of a sense amplifier. 
     By the operation as described above, precharge potential Vpre, i.e., a charging potential accumulated on the decoupling capacitor is made equal to or smaller than potential ext.Vdd (min) which is the lower limit of external supply potential ext.Vdd allowable in terms of specification. Then, even if external supply potential ext.Vdd changes in a precharge operation, a potential higher than the H data potential is never supplied to sense power supply line VSH. Further, the sense amplifier operating voltage generating circuit is operated in the precharge operation to accumulate charges on decoupling capacitor C 10  in the precharge so as to increase gate-source voltage Vgs of a MOS transistor constituting a sense amplifier in the initial stage of a sense amplifier operation. Consequently, a fast sensing operation is possible. 
     Second Embodiment 
     In the first embodiment, precharge potential Vpre is divided by resistors employed in the level shift circuit. 
     However, there arises a need in this case to provide certain adjustment mechanisms respectively for the reference potential generating circuit and the level shift circuit in order to prevent precharge potential Vpre from changing due to process change such as variations in manufacture, and consequently, the size of circuitry increases. In view of this, the sense amplifier operating voltage generating circuit preferably includes only one adjustment mechanism. 
     FIG. 5 is a circuit diagram of a sense amplifier operating voltage generating circuit  600  according to a second embodiment of the present invention. 
     Referring to FIG. 5, a level shift circuit  500  is provided instead of level shift circuit  404  in FIG.  1 . 
     Level shift circuit  500  includes N channel MOS transistors N 61  and N 62  connected in series between an external supply node ext.Vdd and a ground node. The gate of N channel MOS transistor N 61  is supplied with a potential on a sense power supply line VSH while the gate of N channel MOS transistor N 62  is supplied with a reference potential Vref. A shift output potential SVpre is output to a comparator  402  from a node A 60  connecting N channel MOS transistors N 61  and N 62 . 
     Other structural components of the circuitry are the same as those in FIG.  2  and description thereof is not repeated. 
     An operation is described below of sense amplifier operating voltage generating circuit  600  having the circuit structure shown in FIG.  5 . Here, an array operating potential Vdds equal to an H data potential to be written into a memory cell is a constant potential lower than a potential ext.Vdd (min) which is the lower limit of an external supply potential ext.Vdd defined by a specification. 
     It is supposed that reference potential Vref output from a reference potential generating circuit  401  is represented by 
     
       
           Vref =ext. Vdd (min)/2 
       
     
     and that N channel MOS transistors N 61  and N 62  in level shift circuit  500  have the same size and operate in a saturation region. 
     In this case, level shift circuit  500  provides shift output potential SVpre represented by the following equation. 
     
       
           SVpre=Vpre /2 
       
     
     At this time, a P channel driver circuit  403  in a VDC circuit  410  outputs a precharge potential Vpre to sense power supply line VSH in response to variation of external supply potential ext.Vdd as detailed below. 
     (1) External supply potential ext.Vdd is smaller than potential ext.Vdd (min) which is the lower limit of external supply potential ext.Vdd in terms of specification. 
     In this case, precharge potential Vpre supplied from P channel driver  403  is lower than potential ext.Vdd (min). Then, a decreased potential is supplied to the gate of N channel MOS transistor N 61  in level shift circuit  500  and consequently shift output potential SVpre is lower than reference potential Vref (SVpre&lt;Vref). 
     Accordingly, the potential of signal DO output from comparator  402  decreases and a P channel MOS transistor P 40  in P channel driver  403  is constantly made on. 
     Precharge potential Vpre output from P channel driver  403  is thus always equal to external supply potential ext.Vdd and external supply potential ext.Vdd is provided to a decoupling capacitor C 10  in a precharge operation. 
     (2) External supply potential ext.Vdd is higher than potential ext.Vdd (min). 
     In this case, if charging by P channel driver  403  causes precharge potential Vpre to be higher than potential ext.Vdd (min), the gate potential of N channel MOS transistor N 61  in level shift circuit  500  increases and consequently shift output potential SVpre is higher than reference potential Vref (SVpre&gt;Vref). 
     Then, the potential output from comparator  402  decreases and P channel MOS transistor P 40  is turned off. 
     P channel driver circuit  403  is thus controlled to make precharge potential Vpre on sense power supply line VSH equal to potential ext.Vdd (min). 
     By the operation as described above, a relation is established between precharge potential Vpre and external supply potential ext.Vdd as shown by the graph in FIG. 4 like the relation accomplished by sense amplifier operating voltage generating circuit  400  according to the first embodiment. 
     Regarding the sense amplifier operating voltage generating circuit having the structure shown in FIG. 5, adaptation to variation of precharge potential Vpre due to manufacture variations and the like is possible by adjusting reference potential Vref only and no extra adjustment mechanism is necessary. As a result, the size of circuitry can be kept small. 
     Third Embodiment 
     The sense amplifier operating voltage generating circuits according to the first and second embodiments are stopped from operating when signal PRE becomes L level in a sensing operation. Accordingly, in the sensing operation, charges accumulated on the decoupling capacitor are supplied to sense power supply line VSH. 
     In the actual use, external supply potential ext.Vdd could become lower than potential ext.Vdd (min). In this state, precharge potential Vpre is smaller than potential ext.Vdd (min) (Vpre&lt;ext.Vdd (min)). The capacitance of the decoupling capacitor is fixed that is determined by equation (6). Therefore, if external supply potential ext.Vdd becomes lower than potential ext.Vdd (min), the total charge required for a sensing operation cannot be accumulated on the decoupling capacitor. 
     Then, it is desirable that the potential on sense power supply line VSH in a sensing operation can be maintained at array operating potential Vdds equal to H data potential even if external supply potential ext.Vdd is lower than potential ext.Vdd (min). 
     FIG. 6 is a circuit diagram of a sense amplifier operating voltage generating circuit according to a third embodiment of the present invention. 
     Referring to FIG. 6, sense amplifier operating voltage generating circuit  700  includes, instead of reference potential generating circuit  401  shown in FIG. 1, a reference potential generating circuit  701  generating two reference potentials Vref and Vrefs, and an auxiliary VDC circuit  800  is further provided as compared with the circuitry shown in FIG.  1 . Reference potential Vrefs is set to be equal to an array operating potential Vdds. 
     Reference potential generating circuit  701  outputs reference potential Vref to a comparator  402  and outputs reference potential Vrefs to auxiliary VDC circuit  800 . Reference potential generating circuit  701  has the same circuit structure as that of reference potential generating circuit  100  shown in FIG.  10  and description thereof is not repeated. Although reference potentials Vref and Vrefs are generated by the same circuit in FIG. 6, reference potentials Vref and Vrefs may be generated by different circuits respectively. 
     Auxiliary VDC circuit  800  includes a comparator  801  and a P channel driver circuit  802 . 
     Comparator  801  has the same circuit structure as that of comparator  402  and description thereof is not repeated. Comparator  801  receives reference potential Vrefs and precharge potential Vpre to output a signal DO 2  to a P channel MOS transistor P 80  in P channel driver  802 . Comparator  801  further receives a signal SED to operate when signal SED is at H level and stop its operation when signal SED is at L level. 
     Auxiliary VDC circuit  800  is designed to supply a potential equal to array operating potential Vdds to sense power supply line VSH. 
     Although the circuit is structured for reference potential Vrefs equal to array operating potential Vdds (Vrefs=Vdds) in FIG. 6, the relation Vrefs=Vdds is not necessarily required if any appropriate shift circuit is provided in a preceding stage of auxiliary VDC circuit  800 . It is only necessary that a potential provided from auxiliary VDC circuit  800  is equal to array operating potential Vdds. 
     Sense amplifier operating voltage generating circuit  700  having the above circuit structure operates as described below. 
     Here, array operating potential Vdds equal to an H data potential to be written into a memory cell is a constant potential lower than potential ext.Vdd (min) which is the lower limit of external supply potential ext.Vdd in terms of specification. 
     When the capacitance of a decoupling capacitor C 10  is represented by equation (6) and decoupling capacitor C 10  is charged so that potential Vpre on sense power supply line VSH is equal to potential ext.Vdd (min) in a precharge operation, the potential on sense power supply line VSH is always higher than array operating potential Vdds in a sensing operation. Then, charges are never supplied in the sensing operation from auxiliary VDC circuit  800  to sense power supply line VSH. 
     On the other hand, if decoupling capacitor C 10  is charged so that potential Vpre on sense power supply line VSH in a precharge operation is lower than potential ext.Vdd (min), a final potential on sense power supply line VSH in a sensing operation could be lower than array operating potential Vdds. In this case, the operation is controlled to supply charges from auxiliary VDC circuit  800  and make the potential on sense power supply line VSH equal to array operating potential Vdds. 
     In this way, the auxiliary VDC circuit operating in a sensing operation is provided in the sense amplifier operating voltage generating circuit to allow the potential on sense power supply line VSH to be equal to array operating potential Vdds in the sensing operation even if there is a shortage of charges for charging of the decoupling capacitor in a precharge operation due to variation of external supply potential ext.Vdd. 
     When the capacitance of decoupling capacitor C 10  provided in sense amplifier operating voltage generating circuit  700  is smaller than the value determined by equation (6), auxiliary VDC circuit  800  can be operated in a sensing operation to achieve the equal potential on sense power supply line VSH to array operating potential Vdds in the sensing operation. 
     Although the present invention has been described and illustrated in detail, it is clearly understood that the same is by way of illustration and example only and is not to be taken by way of limitation, the spirit and scope of the present invention being limited only by the terms of the appended claims.