Abstract:
The present invention is directed to an improved radar system that produces high range resolution while using existing narrow processing bandwidths and sampling rates to achieve a low cost radar product that is particularly useful for moving targets. The present invention uses a small number of closely spaced Linear Frequency Modulated Chirps. In one embodiment typically 2, 3, 4 chirps are used. Each frequency is sampled at a rate commensurate with the narrower bandwidth, corrected for motion (Time Aligned) and combined to produce a single wide-band chirp but achieved using the lower sample rate commensurate with the narrower transmitted waveform.

Description:
[0001]     This is a conversion of U.S. Provisional Application Ser. No. 60/603,742, filed Aug. 23, 2004, the disclosures of which are incorporated herein by reference. 
     
    
     FIELD OF THE INVENTION  
       [0002]     The present invention relates to improvements in coherent high range resolution imaging applications including but not limited to Strip Map, Spot SAR and ISAR.  
       BACKGROUND OF THE INVENTION  
       [0003]     In many radar applications it is necessary to form two-dimensional images of targets such as ground vehicles, aircraft, ships, and so forth. Resolution in one dimension is provided by the range resolution inherent in the transmit waveform, and resolution in the other cross range dimension is provided by Doppler resolution. The principles are widely applicable and widely applied. Synthetic aperture radar (SAR) forms two-dimensional images in range and cross range with the cross range dimension utilizing the motion of the platform for Doppler resolution. Inverse synthetic aperture radar (ISAR) is accomplishing the same objective with the cross range dimension utilizing the motion of the target being imaged. In other applications the same principle of Doppler resolution is used, even though no specific name has been given to the process. The very same principles also if, instead of forming images, the processor uses range and Doppler resolution to resolve specific scatterers on the target, and then measures the separation of these scatterers in order to obtain target dimensions. All these multi-dimensional imaging techniques require high range resolution (HRR).  
         [0004]     Range resolution (Δr s ) is directly related to bandwidth: 
 
Δ r   s   =c /(2* B ) 
 
 where c is the speed of light and B is the bandwith or frequency excursion of the transmit signal. Modem sub foot imaging systems must therefore transmit and receive an equivalent bandwidth of greater than 600 Mhz. Six inch resolution requires greater than 1.2 Ghz of equivalent bandwidth. The transmitter must be capable of generating this wideband signal and transmitting it without distortion. On receive, the wideband signal must be downconverted and sampled at or above Nyquist for processing. For the six inch case, Nyquist at baseband is 1.2 Ghz. 
 
         [0005]     Direct methods exist that operate with the instantaneous bandwidth being the resolution bandwidth. Two direct waveforms are currently used in obtaining high resolution 1) Direct Short Pulse and 2) Chirped Pulse Compression.  
         [0006]     In direct short pulse systems the time bandwidth product of a rectangular pulse is ˜1. 
 
Bτ˜1 
 
 where τ is the pulsewidth. The range resolution of a direct short pulse is therefore: 
 
Δ r   s =( c *τ)/2 
 
 High range resolution using short pulses is possible with both noncoherent and coherent radars. Magnetron transmitters in noncoherent radars can be turned on and off rapidly enough to generate pulses with a minimum pulsewidth of ˜50 nanoseconds. This has a corresponding range resolution of 24.6 ft. Peak and average power capabilities of a magnetron based system is capable of providing performance at longer ranges however without the resolution. To achieve a range resolution of 1 foot and six inches requires a pulsewidth of ˜2 and 1 nanoseconds respectively. Impulse generators have been used to generate these extremely small pulsewidths. These small pulsewidth can be applied to a High Power Amplifier such as a Traveling Wave Tube Amplifier (TWTA), however even with the higher peak power, the average powers achievable limits the long range performance. For this reason the sub foot direct pulse configurations have been primarily used in Radar Cross Section diagnosis&#39;s configurations. 
 
         [0007]     Chirped pulse compression allows a radar system to transmit a pulse of relatively long duration pulse (microseconds) at a higher peak power pulse to simultaneously attain the range resolution of a short pulse and the high average power of a long pulse transmitter. Fine range resolution is achieved in pulse compression systems by coding the RF carrier of the long pulse to increase the bandwidth of the transmit signal. When the reflected signal is received, the coded waveform is applied to a matched filter that compresses the transmitted pulse. Most airborne radars use linear Frequency Modulation (LFM) where the frequency of the transmit signal varies linearly with time. For example, the transmit frequency of one radar varies by 185 MHz over a 40 microsecond pulse. The receiver collects reflected pulses, which are 3.25 nm long and compresses them into an effective pulse width that is 5.4 nanoseconds. This corresponds to a resolution of 0.8 m. The 10 kilowatt peak power transmitter of the transmitter provides an average power of 200 watts on targets out to 160 nm. If a magnetron-based radar system could transmit a 5.4 nanosecond pulse, it would have to output 74 MW of power to achieve the same average power on target. It would not be practical to fly such a transmitter on a aircraft because of the larger size, prime power consumption and cooling apparatus. As a result of pulse compression, the 25 foot resolutions achieved with magnetron systems have been reduced to less than 1 m. Chirped pulse compression systems can achieve sub foot resolutions by transmitting and receiving the necessary bandwidth in the coded waveform. However, as mentioned previously, the transmit and receive hardware must support the instantaneous bandwidth. At X-Band the ratio of bandwidth to carrier frequency to achieve sub foot resolution is &gt;10%. This pushes the hardware up against technology constraints and results in an expensive and complex radar system.  
         [0008]     Stretch processing of linear FM pulses originally developed by W. Caputi, reduced the complexity on the receiver side of the radar. The stretch concept provides HRR by transmitting a linear FM pulse of the necessary bandwidth just like a chirped pulse compression system. On receive the returns are down-converted in frequency with a frequency modulated Local Oscillator signal of identical FM slope as the transmit signal. The corresponding down-converted signals are fixed frequency pulses, the center frequency of the pulse depends upon the relative range of the range gated scene. The IF signals are sampled and converted to the frequency domain providing a range profile of the scene. This technique does reduce the complexity of the receive path however the transmit path remains complex and the technique is not extendable to larger range swaths.  
         [0009]     To bypass the complexity, cost and technological limitations of achieving subfoot high range resolution with transmit signals containing the necessary instantaneous bandwidth, engineers have developed a technique to synthesize the HRR using multiple narrower band signals of different frequencies. This technique, known as the stepped frequency waveform (SFW) consists of a sequence of pulses transmitted with fixed uniform pulse to pulse frequency change (ΔF). The number of pulses (N) required is a function of the desired resolution and the pulse to pulse frequency change (ΔF). 
 
Δ r   s   =c /(2* N*ΔF ) 
 
 The SFW process is not a single look process like the direct methods described above. It requires the transmission and reception of multiple pulses. The total time to transmit the necessary pulses to synthesize the HRR waveform is simply the radar PRI times the number of pulses. Along with the simplicity of the technique there are many limitations that limit its usefulness. Conventional Step Frequency processing does not handle moving targets either actual motion or motion due to scene rotation about a fixed point in a Spot SAR or ISAR image. It requires a large number of individual frequencies to permit acceptable sample weighting to achieve low range sidelobes after the DFT. The coarse range samples are unmodulated pulses. The large number of frequencies required may mean that there are insufficient samples to permit target Doppler Frequency measurement and there may be a multitude of adjacent channel self-jamming situations. DFT/FFT processing requires a constant time step between transmissions. This will exacerbate any target motion 
 
       SUMMARY OF THE INVENTION  
       [0010]     The present invention produces high range resolution while using existing narrow processing bandwidths and sampling rates to achieve a low cost radar product that is particularly useful for moving targets. The present invention uses a small number of closely spaced Linear Frequency Modulated Chirps. In a preferred embodiment typically 2, 3, 4 or more chirps are used. Each frequency is sampled at a rate commensurate with the narrower bandwidth, corrected for motion (Time Aligned) and combined to produce a single wide-band chirp but achieved using the lower sample rate commensurate with the narrower transmitted waveform. For a moving Radar platform the samples themselves are also corrected for platform motion. Following combination, the new wide band waveform is pulse compressed in the conventional manner. The method allows the individual frequencies to be transmitted with arbitrary time separation. By transmitting the pulses with small time separation there are fewer target motion effects and more accurate corrections. 
     
    
     DESCRIPTION OF THE FIGURES  
       [0011]      FIG. 1  is an example of a long range case Time Aligned Burst Step (TABS) Frequency.  
         [0012]      FIG. 2  is an example of a short range case TABS Frequency.  
         [0013]      FIG. 3  is a conceptual signal flow of the present invention.  
         [0014]      FIG. 4  is a block diagram of the step frequency processing.  
         [0015]      FIG. 5  is a block diagram of the time alignment.  
         [0016]      FIG. 6  is a signal plot of the matched filter.  
         [0017]      FIG. 7  is a graph of the matched filtered frequencies blended/concatenated with a 5 MHz overlap.  
         [0018]      FIG. 8  shows a graph of the range compress.  
         [0019]      FIG. 9  shows the Chi-Square fit to Reference IPR.  
         [0020]      FIG. 10  illustrates one example of the invention: 2×0 m/s, TA disabled.  
         [0021]      FIG. 11  illustrates a second example of the invention: 2×0 m/s, TA enabled.  
         [0022]      FIG. 12  illustrates a third example of the invention: 3 &amp; 0 m/s, TA disabled.  
         [0023]      FIG. 13  illustrates a fourth example of the invention: 3 &amp; 0 m/s, TA enabled.  
         [0024]      FIG. 14  illustrates a fifth example of the invention: 0 &amp; 0 m/s, TA disabled.  
         [0025]      FIG. 15  illustrates a sixth example of the invention: 0 &amp; 0 m/s, TA enabled.  
         [0026]      FIG. 16  illustrates a seventh example of the invention: 3 &amp; 0 m/s, TA disabled.  
         [0027]      FIG. 17  illustrates an eighth example of the invention: 3 &amp; 0 m/s, TA enabled. 
     
    
     DETAILED DESCRIPTION OF THE INVENTION  
       [0028]     In the present invention a transmitter is employed to broadcast a frequency-modulated probe signal at each of a number of frequency steps. A receiver receives a return signal from which magnitude and phase information corresponding to a target object are measured and stored in a memory at each of the center frequency steps. The range to the object is determined using the set of magnitude and phase information stored in the memory. The present invention uses a number of narrow bandwidth pulses instead of a large broad band pulse to determine the location of a target.  
         [0029]      FIG. 1  illustrates the present invention being used to locate a target at a long range. The pulses F 1 , F 2 , and F 3  represent a series of pulses emitted by a radar system utilizing the present invention. The pulses R 1 , R 2 , and R 3  represent the pulses returning to the detector after bouncing off the target. All the F pulses in the initial emission are transmitted in a series of short bursts before any of the R pulses have returned from the target. The number of F pulses in each emission can vary in the period. The greater the number of chirps that are transmitted before the R pulses return, the less target motion effects that are present. The transmission of the F pulses prior to return of the R pulses produces high range resolution while using narrow processing bandwidths. The F pulses are preferably closely spaced Linear Frequency Modulated Chirps. Each frequency is sampled at a rate commensurate with the narrow bandwidth, corrected for motion and combined to produce a single wide band chirp but obtained from the lower sample rate commensurate with the narrower transmitted waveform.  
         [0030]      FIG. 2  illustrates the present invention being used to locate a target at a short range. The pulses F 1 , F 2 , and F 3  represent a series of pulses emitted by a radar system employing the present invention. The pulses R 1 , R 2 , and R 3 , represent the pulses returning to the detector after bouncing off the target. Notice that the corresponding R pulse for each F pulse has bounced off the target and returned to the detector before the next F pulse is sent.  
         [0031]      FIG. 3  illustrates a block diagram of one embodiment of the present invention. The Time Aligned Burst Step-Frequency Radar Front End,  31  transmits, for example, 3 signals designated as, F 1  ( 32 ), F 2  ( 33 ), and F 3  ( 34 ). Signal F 1  ( 32 ) and F 2  ( 33 ) are sent to the Coarse Motion Compensation  35 , while F 3  ( 34 ) is sent to Time Alignment  36 . The three signals are then combined to form a single signal  37  and sent to Pulse Compression  38 . Pulse Compression creates the final High Range Resolution Signal  39 .  
         [0032]      FIG. 4  illustrates a block diagram of another embodiment of the present invention. The two swaths,  41  and  42 , are sent to the Time Align. The time aligned signal is then sent to a frequency grid shift  44  where a third swatch  43  is convolved with the time aligned signal. The Frequency Grid shift signal is then sent to the Pre-calculated matched filter  46 . The signal from the Matched Filter  46  is then sent to the Blend  47  and is finally sent to the Range Compress  48 . The Range Compressed signal  49  is the final product.  
         [0033]      FIG. 5  illustrates a block diagram showing the internal workings of the Time Align  45  block in  FIG. 4 . If pulse trains of different frequencies are transmitted at different times, the target may move appreciably in phase during this time. This effect can be minimized if the system interpolates the pulses in slow time back to the pulse time center frequency. The resolution is assisted by the fact that the delays can be minimized, approximately 75 microsec. This linear interpolation is sufficient.  
         [0034]      FIG. 6  illustrates the Matched filter  46  in  FIG. 4 .  FIG. 6  is precalculated as the conjugate 16K:FFT of an idealized chirp (205 MHz BW20 microsec pulse width). This is applied individually to each swath. The swath is Fourier interpolated to correct the frequency grid by applying a phase ramp to the time chirp. The output for point targets should be linear ramp matching its neighboring swaths.  
         [0035]      FIG. 7  illustrates the Blend  47  block in  FIG. 4 . The Match Filtered frequencies are blended/concatenated with a 5 MHz overlap. There is linear weighting for a coherent sum.  
         [0036]      FIG. 8  illustrates a wave form of the Range Compress  48  block shown in  FIG. 4 . Pad for 65 k IFFT, Hamming window for side lobe reduction. The final output is 15 meter sampling, 0.25 meter resolution two target case at −1000 meters and 1000 meters.  
         [0037]     It is assumed that the hamming window is 31% wider than sinc, to achieve 1 ft resolution Bw=645 Mhz. The predicted resolution for 605 MHz is 0.32 m. The firing sequence time offsets has no effect on stationary targets, and superposition of targets hold.  
         [0038]     With reference to  FIG. 9  the graph may be used to estimate resolution and position offset in non-grid centered target responses. The graph also permits estimates of PSLR and ISLR to be made. The graph shows sample indexes with 0.15 m sampling resolution.  
         [0039]      FIG. 10 - FIG. 17  demonstrate the time alignment improvement for the following: 
    For targets separated in range     For targets separated at the same range and     For targets at different velocities.      
         [0043]     The parameters are shown in Table 1 as follows  
                                                                             Case/                           Sce-   Time   Range (m)   Range (m)   Range (m/s)   Range (m/s)       nario   Alignment   Target #1   Target #2   Target #1   Target #2                                1   Disabled   −1000   +1000   0   0       2   Enabled   −1000   +1000   0   0       3   Disabled   −1000   +1000   0   3       4   Enabled   −1000   +1000   0   3       5   Disabled   0   0   0   0       6   Enabled   0   0   0   0       7   Disabled   0   0   0   3       8   Enabled   0   0   0   3                  
 
 Case/Scenario Parameters 
    Scenario #1: 2 &amp; 0 m/s, TA disabled    
 
         [0045]     Target at −1 km: 3 dB=0.322 m, PSLR=−41.8 dB, ISLR=−35.6 dB  
         [0046]     Target at −1 km: 3 dB=0.322 m, PSLR=−41.9 dB, ISLR=−35.6 dB 
    Scenario #2: 2 &amp; 0 m/s, TA enabled    
 
         [0048]     Target at −1 km: 3 dB=0.322 m, PSLR=−41.8 dB, ISLR=−35.6 dB  
         [0049]     Target at −1 km: 3 dB=0.322 m, PSLR=−41.9 dB, ISLR=−35.6 dB 
    Scenario #3: 3 &amp; 0 m/s, TA disabled    
 
         [0051]     Target at −1 km, 3 m/s: 3 dB=0.322 m, PSLR=−34.6 dB, ISLR=−30.7 dB  
         [0052]     Target at −1 km, 0 m/s: 3 dB=0.322 m, PSLR=−42.1 dB, ISLR=−35.7 dB 
    Scenario #4: 3 &amp; 0 m/s, TA enabled    
 
         [0054]     Target at −1 km, 3 m/s: 3 dB=0.327 m, PSLR=−40.7 dB, ISLR=−35.0 dB  
         [0055]     Target at −1 km, 0 m/s: 3 dB=0.322 m, PSLR=−42.1 dB, ISLR=−35.7 dB 
    Scenario #5: 0 &amp; 0 m/s, TA disabled    
 
         [0057]     Two targets at 1 km: 3 dB=0.322 m, PSLR=−42.0 dB, ISLR=−35.6 dB 
    Scenario #6: 0 &amp; 0 m/s, TA enabled    
 
         [0059]     Two targets at 1 km: 3 dB=0.322 m, PSLR−42.0 dB, ISLR=−35.6 dB 
    Scenario #7: 3 &amp; 0 m/s, TA disabled    
 
         [0061]     Targets at 1 km, 0 &amp; 3 m/s: 3 dB=0.323 m, PSLR=−36.7 dB, ISLR=−31.0 dB 
    Scenario #8: 3 &amp; 0 m/s, TA enabled    
 
         [0063]     Targets at 1 km, 0 &amp; 3 m/s: 3 dB=0.325 m, PSLR=−40.4 dB, ISLR=−34.3 dB