Abstract:
An output circuit is provided which exhibits a waveform having a higher edge rate, with less ringing and power consumption than many conventional differential amplifier output driver circuits. A pre-driver stage using a current-mode logic (CML) design eliminates the frequency dependent transfer characteristics associated with emitter follower amplifiers used with emitter-coupled logic (ECL)-pre-drivers. The final stage CML circuit has been modified to eliminate the Miller-effect capacitance, using cascode transistors to maintain a constant voltage at the collectors of the final stage CML circuit transistors. The cascode transistors isolate the switching noise of the final stage CML transistor pair. Further, the bases of the final stage CML transistors present a smaller load to the pre-driver stage output, permitting the pre-driver stage to be a CML rather than an ECL design. A method of amplifying a differential signal in accordance with the principles of the above-described circuit is also provided.

Description:
BACKGROUND OF THE INVENTION 
     The invention relates generally to low impedance output stage driver circuits and, more particularly, to a modification of a current-mode logic (CML) circuit which permits high fidelity amplification of high speed signals into low impedance loads using an output stage differential CML amplifier. 
     It is well known to use a so-called differential amplifier to amplify a differential signal, such as a signal that may be used in a high speed transfer of digital data. When bipolar junction n-p-n transistors are used, such an amplifier is made by tying the emitters of two transistors together. The differential input signals are applied to the bases of the two transistors, and amplified differential output signals are provided at the transistor collectors. This type of differential amplifier is called CML circuit. 
     CML circuits are often used to amplify differential digital signals, where the first digital signal is the complement of the second digital signal. A constant current source is connected to coupled emitters, and relatively small changes to the emitter-coupled transistor base currents cause the constant current to flow from one of the emitter-coupled transistors, to the other. That is, the output voltages rapidly change from high to low, or visa versa. A digital type on/off signal that is input into such a circuit will be amplified, with a change in the dc level. 
     The dc level change in the amplified signal is typically not desired, and these dc levels can be shifted using emitter follower amplifiers. Besides providing a dc level shift, this configuration of the differential amplifier, called an emitter-coupled logic (ECL) circuit, also provide a greater drive capacity to subsequently connected loads. However, the greater drive comes at the expense of frequency dependent amplifier impedances which causes infidelities in signal amplification, such as overshoot and ringing. To some extent the frequency dependent transfer characteristics of the transistor can be mitigated by operating the emitter followers at higher current levels. However, in integrated circuit (IC) design power consumption is critical. 
     FIG. 1 is a schematic diagram of an output CML circuit using ECL and CML circuits (prior art). The output circuit  10  comprises a pre-driver stage  12  and a final stage  14 . The pre-driver stage  12  is an ECL circuit as described above, where Q 3  and Q 4  are emitter-coupled transistors, and Q 1  and Q 6  are emitter followers used to interface the pre-driver stage  12  with the final stage  14 . Q 5  and R 4  act as a constant current source, while Q 2 /R 1  and Q 7 /R 5  are used to bias the emitter followers Q 1  and Q 6 . The differential inputs A and A 1  are connected, respectively, to the bases of Q 3  and Q 4  on lines  16  and  18 . The pre-driver outputs N 2  and N 1  are on lines  20  and  22 . 
     The final stage  14  is a CML circuit as described above. Q 8  and Q 9  are emitter-coupled transistors, while Q 1 /R 8  acts as a constant current source. The inputs N 2  and N 1  are connected on lines  20  and  22 , respectively with the bases of Q 9  and Q 8 . The final stage outputs Y and Yn are connected on lines  24  and  26 , respectively to Q 9  and Q 8 . When the circuit of FIG. 1 is an output driver circuit to drive large loads, the resistances of R 6  and R 7  are low, for example, 50 ohms. 
     FIG. 2 is an exemplary signal diagram illustrating signal degradation in the amplification process (prior art). Signal A on line  16  is shown as an ideal digital signal with near-perfect rise and fall times, and no overshoot or excessive damping characteristics. Signal N 1  is the amplified signal on line  22  that is normalized with respect to gain. Alternately, signal N 2  could be displayed having a polarity opposite to the A signal. Amplification has introduced imperfect transitions. These imperfections are compounded in the next stage of amplification as shown in signal Y, which has also been normalized with respect to gain. Overshoot can be mitigated by biasing emitter follower transistors Q 1  and Q 6  to operate at a higher quiescent current level. 
     Slow rise and fall times seen at the transitions of signals N 1  and Y of FIG. 2 are largely due to the so-called Miller-effect capacitance. The Miller-effect capacitance acts to vary to input capacitance from the base of a transistor to the collector in response to the voltage presented to the base, making the transistor input impedance vary with respect to the frequency of the input signal. As the voltage on the base increases, for example on Q 8 , the signal is amplified and the voltage on the collector simultaneously decreases. The Miller-effect capacitance causes an increased parasitic current flow from base to collector, in a sense acting as a larger capacitor, and taking away current that would otherwise flow into the base of Q 8 . 
     The overshoot and ringing seen at the transitions of signal Y of FIG. 2 are largely due to the use of emitter follower amplifiers Q 1  and Q 6  (FIG.  1 ). As mentioned above, the emitter follows are used to shift the dc level of the output signal and to increase high frequency gain between stages. However, the improved high frequency response comes at the price of frequency dependent gain that promotes ringing. 
     It is known to use feedback capacitors between the base and collector of a transistor to improve the flatten the frequency response of an amplifier. To some extent this feedback minimizes the signal degradation problems associated with the circuit of FIG.  1 . Cross-coupled feedback capacitors have been used in differential amplifiers. For example, a capacitor from the base Q 8  to the collector of Q 9 , and a capacitor from the base of Q 9  to the collector of Q 8 . However, these solution typically come at the expense of diminished frequency response. It is also known to form a cascode transistor combination to form an amplifier with improved frequency response and less sensitivity to Miller-effect capacitance. Many high-speed communication processes require circuitry that generates clean waveforms, promoting quicker recognition to changes in state and more resistance to error in the transfer of data. Improvements in the waveforms produced by differential amplifier driver output circuitry are required to support the above-mentioned circuits. 
     It would be advantageous if the Miller-effect capacitance associated with the bases of emitter-coupled transistors in an output circuit CML differential amplifier could be eliminated. 
     It would be advantageous if CML stages could be coupled without the necessity of emitter follower amplifiers to provide cleaner output signal transitions with less overshoot and ringing. 
     It would be advantageous if signal integrity in the amplification of signals by a differential amplifier could be improved without increasing the operating currents of the drive circuitry. 
     SUMMARY OF THE INVENTION 
     Accordingly, an output driver circuit is provided comprising a final stage differential amplifier including emitter-coupled first and second transistors, and a pre-driver stage differential amplifier including emitter-coupled first and second transistors. A cascode is connected to the final stage differential amplifier to maintain a constant voltage at the collectors of the first and second transistors of the final stage differential amplifier. 
     The cascode is a pair of transistors cascoded with the differential amplifier. A first cascode transistor has a collector connected to Vcc through a first load resistor and an emitter connected to the collector of the final stage first transistor. A second cascode transistor has a collector connected to Vcc through a second load resistor and an emitter connected to the collector of the final stage second transistor. 
     The cascode also includes a first current bleeder transistor having a collector connected to the collector of the final stage first transistor and an emitter operatively connected to ground, and second current bleeder transistor having a collector connected to the collector of the final stage second transistor and an emitter operatively connected to ground. 
     A method is also provided for amplifying differential signals, the method comprising: 
     receiving a pair of differential input signals at a corresponding pair of circuit inputs; 
     differentially amplifying the voltage of the differential input signals; 
     simultaneously with the amplification of the differential input signals, eliminating any changes in the capacitance of the circuit inputs responsive to the amplification of the differential input signals; and 
     providing a pair of differential output signals which are amplified replicas of the corresponding differential input signals. 
     As in the above description of the circuit, the elimination of capacitance changes at the circuit inputs due to signal-amplification includes eliminating the Miller-effect capacitance at the bases of the final stage emitter-coupled transistors. The elimination of capacitance changes is accomplished in two sub-steps, comprising: 
     maintaining. a constant voltage at the collectors of the final stage emitter-coupled transistors; and. 
     bleeding current from the collectors of the final stage emitter-coupled transistors. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is. a schematic diagram of an output CML circuit using ECL and CML circuits (prior art). 
     FIG. 2 is an exemplary signal diagram illustrating signal degradation in the amplification process (prior art). 
     FIG. 3 is a schematic diagram illustrating the output driver circuit of the present invention. 
     FIGS. 4 a  and  4   b  are more detailed depictions of the schematic drawing of FIG.  3 . 
     FIG. 5 is a flowchart illustrating the present invention method for amplifying differential signals. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     FIG. 3 is a schematic diagram illustrating the output driver circuit of the present invention. The output driver circuit  100  comprises a final stage differential amplifier  102 , including emitter-coupled first and second transistors Q 104  and Q 106 , respectively. Final stage differential amplifier  102  includes a pair of differential outputs Yn and Y on lines  108  and  110 , respectively, and a pair of differential inputs N 1  and N 2  on lines  112  and  114 , respectively. 
     Output driver circuit  100  also comprises a pre-driver stage differential amplifier  116 , including emitter-,coupled first and second transistors Q 118  and Q 120 , respectively. The pre-driver differential amplifier  116  also includes a pair of differential outputs N 1  and N 2  connected to the differential inputs of the final stage differential amplifier  102  on lines  112  and  114 . The pre-driver differential amplifier  116  has differential pre-driver input signals A and An on lines  122  and  124 , respectively. 
     A cascode  128  is connected to the final stage differential amplifier  102  to maintain a constant voltage at the collectors (c) of the first and second transistors Q 104  and A 106  of the final stage differential amplifier  102 . 
     A first load resistor R 130  has a first input operatively connected to the collector (c) of the final stage differential amplifier first transistor Q 104  through the cascode  128 . The second input to the first load resistor R 130  is connected to Vcc. A second load resistor R 132  has a first input operatively connected to the collector of the final stage differential amplifier second transistor Q 106  through the cascode  128 . Typically, the first load resistor R 130  and the second load resistor R 132  are 50 ohms, but any resistance value can be used. Collectively, R 130  and R 132  are an example of an output impedance means  133 . 
     Alternately stated, output driver circuit  100  comprises a final stage means  102  for differentially amplifying a pair of input signals (N 1  and N 2 ) on lines  112  and  114 . A pair of output signals (Yn and Y) are generated on lines  108  and  110 . A pre-driver stage means  116  for differentially amplifying a pair of pre-driver input signals (A and A 1 ) on lines  122  and . 124  provides the input signals N 1  and N 2  to the final stage differential amplifying means  102 . 
     The circuit  100  also includes a means for buffering  128  the output signals on lines  108  and  110  of the final stage differential amplifying means  102  from changes to the input impedance of the final stage differential amplifying means  102  on lines  112  and  114 . 
     The final stage differential amplifying means  102  includes the first and second emitter-coupled transistor Q 104  and Q 106 , and the output impedance means  133  includes a first and second load resistors R 130  and  132  in some aspects of the invention. 
     FIGS. 4 a  and  4   b  are more detailed depictions of the schematic drawing of FIG.  3 . All the circuit elements are shown in FIG. 4 a , while FIG. 4 b  specifically illustrates the cascode  128 . The cascode  128  includes a pair of cascoded transistors Q 134  and Q 136 . The first cascode transistor Q 134  has a collector connected to the first load resistor R 130  first input and an emitter connected to the collector of the final stage differential amplifier first transistor Q 104 . The second cascode transistor Q 136  has a collector connected to the second load resistor R 132  first input and an emitter connected to the collector of the final stage differential amplifier second transistor Q 106 . 
     The cascode  128  also includes a first current bleeder  138  (see FIG. 4 b ) connected to the collector of the final stage differential amplifier first transistor Q 104 , and a second current bleeder  140  connected to the collector of the final stage differential amplifier second transistor Q 106 . The current bleeders  138  and  140  continuously sink current so that when either Q 104  or Q 106  is not “on”, their collector voltages will remain constant. The first current bleeder  138  includes a first current bleeder transistor Q 142  having a collector connected to the collector of the final stage differential amplifier first transistor Q 104 . The second current bleeder  140  includes a second current bleeder transistor Q 144  having a collector connected to the collector of the final stage differential amplifier second transistor Q 106 . 
     Returning to FIG. 4 a , the collector of the pre-driver differential amplifier first transistor Q 118  is connected to the base of the final stage differential amplifier second transistor Q 106  (N 2 ) on line  114 . The collector of the pre-driver differential amplifier second transistor Q 120  (N 1 ) is connected to the base of the final stage differential amplifier first transistor Q 104  on line  112 . 
     The cascode  128  includes a bias circuit for the first and second cascode transistors Q 134  and Q 136 . A first bias transistor Q 146  has an emitter connected to the bases the first and second cascode transistors Q 134  and Q 136 . A second bias transistor Q 148  has a collector connected to the base of the first bias transistor Q 146 . A third bias transistor Q 150  has a collector connected to the emitter of the first bias transistor Q 146  and a base connected to the base of the second bias transistor Q 148 . The base of bleeder transistors Q 142  and Q 144  are also connected to the base of the second bias transistor Q 148 . 
     It should be noted that the bias circuits described above, and shown in FIG. 4 a , are only some of many possible and widely known techniques that are available to enable the above-mentioned differential amplifier, cascode, current source, and bleeder transistors. Generally, a bias system is designed to provide consistent amplifier performance at all levels of signal amplification, across wide temperature variations, and across different part tolerance and performance variations. Other bias schemes can be used to enable the present invention,especially if specific amplifier performance features are desired. 
     Alternately stated, the buffering means  128  includes a pair of cascoded transistors Q 134  and Q 136 . The buffering means  128  also includes a first means for bleeding current  138  (see FIG. 4 b ) connected to the collector of the final stage first transistor, and a second means for bleeding current  140  connected to the collector of the final stage second transistor  22 . The first current bleeding means  138  includes a first current bleeder transistor Q 142  having a collector connected to the collector of the final stage first transistor Q 104  and second current bleeder transistor Q 144  having a collector connected to the collector of the final stage second transistor Q 106 . 
     Returning to FIG. 4 a , the pre-driver differential amplifying means  116  includes the first transistor Q 118  with a collector connected to the base of the final stage second transistor Q 106 , and the second transistor Q 120  with a collector connected to the base of the final stage first transistor Q 104 . 
     To power the output driver circuit  100  a first reference voltage having a first voltage is provided. The first reference voltage is depicted in FIG. 4 a  as Vcc. A second reference voltage, having a second voltage less than the first voltage is also included. The second reference voltage is depicted as ground. 
     A first bias resistor R 156  has a first input connected to the base of the first bias transistor Q 146  and a second input connected to the first reference voltage (Vcc). A second bias resistor R 158  has a first input connected to the emitter of the second bias transistor Q 148  and a second input connected to the second reference voltage (gnd). A third bias resistor R 160  has a first input connected to the emitter of the third bias transistor Q 150  and a second input connected to the second reference voltage. 
     A first current bleeder resistor R 162  has a first input connected to the emitter of the first current bleeder transistor Q 142  and a second input connected to the second reference voltage. A second current bleeder resistor R 164  has a first input connected to the emitter of the second current bleeder transistor Q 144  and a second input connected to the second reference voltage. 
     A first constant current resistor R 166  has a first input connected to the emitter of the first constant current transistor Q 152  and a second input connected to the second reference voltage. A second constant current resistor R 168  has a first input connected to the emitter of the second constant current transistor Q 154  and a second input connected to the second reference voltage. 
     A tap resistor R 170  has a first input connected to the first reference voltage and a second input connected as described below. A first collector resistor R 172  has a first input connected to the second input of the tap resistor R 170 , and a second input connected to the collector of the pre-driver stage differential amplifier first transistor Q 118 . A second collector resistor R 174  has a first input connected to the second input of the tap resistor R 170 , and a second input connected to the collector of the pre-driver stage differential amplifier second transistor Q 120 . 
     Further, the collector of the first bias transistor Q 146  is connected to the first reference voltage, and the first and second load resistors R 130  and R 132  have second inputs connected to the first reference voltage. 
     The final stage differential amplifier  102  has a first .differential output (Yn) connected to the first input of the first load resistor R 130  on line  108 , and a second differential output (Y) connected to the first input of the second load resistor-R 132  on line  110 . 
     FIG. 5 is a flowchart illustrating the present invention method for amplifying differential signals. Although the method is described as a series of numbered steps for the purpose of clarity, no order should be inferred from the numbering, unless explicitly stated. Step  200  provides a circuit having inputs. In some aspects of the invention these circuit inputs can be described as voltage inputs. Step  202  receives a pair of differential input signals at a corresponding pair of circuit inputs. In some aspects of the invention the input signals can be described as voltage input signals. Step  204  differentially amplifies the differential input signals. In some aspects of the invention Step.  204  describes a voltage gain. Simultaneously with the amplification of the differential input signals in Step  204 , Step  206  eliminates any changes in the capacitance of the circuit inputs responsive to the amplification of the differential input signals. Alternately stated, Step  206  maintains a consistent gain characteristic across the range of input signal frequencies. Step  208  is a product, providing a pair of differential output signals which are amplified replicas of the corresponding differential input signals. 
     In some aspects of the invention Step  200  provides a final stage differential amplifier, including an emitter-coupled transistor pair. The receiving of the differential input signals in Step  202  includes receiving a varying voltage input signal at each of the bases of the final stage emitter-coupled transistors. The elimination of capacitance changes at the circuit inputs due to amplification in Step  206  includes eliminating the Miller-effect capacitance it the bases of the final stage emitter-coupled transistors as the differential input signals vary in voltage. 
     The elimination of capacitance changes at the circuit inputs due to amplification in Step  206  includes sub-steps. Step  206   a  maintains a constant voltage at the collectors of the final stage emitter-coupled transistors. In some aspects of the invention Step.  200  provides a cascode, including a pair of transistors cascoded with the final stage emitter-coupled transistors. The elimination of capacitance changes at the circuit inputs due to amplification in Step  206  includes using the cascode transistor pair to maintain a constant voltage at the collectors of the final stage emitter-coupled transistors in Step  206   a.    
     The elimination of capacitance changes at the circuit inputs due to amplification in Step  206  includes a second sub-step. Step  206   b  bleeds current from the collectors of the final stage emitter-coupled transistors. 
     In some aspects of the invention Step  200  provides a pre-driver stage differential amplifier including a pair of emitter-coupled transistors. Then, Step  201  (not shown) provides the differential input signal from corresponding collectors of the pre-driver differential amplifier emitter-coupled transistors. 
     The circuit described above in FIGS. 3,  4   a , and  4   b , and the method described in FIG. 5, are enabled using n-p-n transistors and a first reference voltage having a higher potential than the second reference voltage. Alternately, an equivalently performing circuit could be embodied using p-n-p transistors, combinations of n-p-n and p-n-p transistors, and alternately referenced voltages. Such alternate embodiments have not been specifically described herein as one reasonably skilled in the art would be able to build these circuits from the explanations of FIGS. 3,  4   a , and  4   b.    
     Another embodiment of the present invention concept would use differential amplifiers of source-coupled field-effect transistors (FET)s, or the like. FET transistors also have the problem of Miller-effect capacitance on the gates of the transistors that results in frequency dependent amplifier gain. The invention could be embodied using FET differential amplifiers and bipolar cascode transistors, bipolar differential amplifiers and FET cascode transistors, or all FET transistors. Likewise, the transistor combinations can be varied between pre-driver and final stages. However configured, the voltage at the drain of the FET differential amplifier transistor would be maintained at a near constant level despite the input signal. These, and other embodiments and variations of the above-described invention will occur to those skilled in the art.