Abstract:
Phase noise analyzing system that analyzes phase noise in radio frequency (RF) signals provided by a device under test when coupled thereto includes a low noise receiver that receives at a first input, signal from the device under test when coupled thereto, a low noise synthesizer that provides output to the receiver, a frequency domain analyzer that receives output from the receiver, a time domain analyzer that receives output from the receiver, a switching system that controls signal flow to and from the receiver, the synthesizer, the frequency domain analyzer and the time domain analyzer, and a computer that controls the switching system to perform the analysis of phase noise in signals provided by the device under test.

Description:
FIELD OF THE INVENTION 
     The present invention relates generally to the testing/screening of radar system components at the flight line level for phase noise performance and to methods and systems that perform absolute, additive, AM, PM, pulsed and direct phase noise measurements of radio frequency (RF) signals. 
     BACKGROUND OF THE INVENTION 
     Radar systems with Built-In Test (BIT) target generators typically provide only limited target detection test capability. Excessive phase noise within the Stable Local Oscillator (STALO) of a radar system can significantly impact target detection in pulse Doppler and moving target indicator radars. Once a radar system leaves the factory, it typically is not screened for phase noise ever again. As components within the radar system degrade, they can affect phase noise and overall system performance, often adversely, without exhibiting functional failures or any indication to an operator. 
     OBJECTS AND SUMMARY OF THE INVENTION 
     An object of at least one embodiment of the present invention is to provide aircraft maintenance personnel with a test system that permits periodic screening of radar system assemblies for phase noise on the actual aircraft, and importantly, before component degradation significantly affects performance. 
     A phase noise analyzing system that analyzes phase noise in radio frequency (RF) signals produced by a device under test when coupled thereto in accordance with the invention includes a low noise receiver that receives at a first input, signal from the device under test when coupled thereto, a low noise synthesizer that provides output to the receiver, a frequency domain analyzer that receives output from the receiver, a time domain analyzer that receives output from the receiver, a switching system that controls signal flow to and from the receiver, the synthesizer, the frequency domain analyzer and the time domain analyzer, and a computer that controls the switching system to perform the analysis of phase noise in signals provided by the device under test. 
     The computer may be a portable computer, in which case, a transportable housing has a first port for connecting to the computer and a second port for connecting to the device under test, and the receiver, the synthesizer, the frequency domain analyzer and the time domain analyzer are situated in or on the housing such that the system is transportable. Also, a power supply maybe arranged in or on the housing and coupled to the receiver, the synthesizer, the frequency domain analyzer and the time domain analyzer to provide power thereto. 
     Advantageously, for the transportable system, a power supply is arranged in or on the housing and configured to provide power to the device under test when coupled thereto. 
     A method for analyzing phase noise in signals provided by a device under test in accordance with the invention includes measuring phase noise of a signal from the device under test, setting, using a processor, an initial corrected value for points in the signal to a value (MV) of the measured phase noise and a counter to 1, then computing, using a processor, a corrected value (CV) for each point based on the equation 
             CV   =     MV   -     10   *       log   10     ⁡     (     1   +     10     -     ABS   ⁡     [       REF   -   CV     10     ]             )                 
wherein REF is a level of a reference carrier of the low noise synthesizer, and determining factor for each point, using a processor, whether a change between the corrected value and an immediately preceding corrected value is below a threshold or whether the counter is greater than a predetermined number and if so, using the corrected value as an indication of the phase noise of the signal, otherwise incrementing the counter and re-computing the corrected value until the change between the corrected value and an immediately preceding corrected value is below the threshold or the counter is greater than the predetermined number. This may be referred to as software correction for tangential.
 
     Another method for analyzing phase noise in signals provided by a device under test in accordance with the invention includes measuring phase noise of a signal from the device under test using a receiver having a phase lock loop with a voltage controlled crystal oscillator that induces a suppression effect proximate a carrier frequency of the signal, for a frequency range of the signal, determining a polynomial that best fits points of the signal in the frequency range, and increasing the measured phase noise by a line determined by the polynomial fit to correct for the suppression induced by the phase lock loop. This may be referred to as software correction for loop bandwidth correction. 
     Yet another method for analyzing phase noise in signals provided by a device under test in accordance with the invention includes identifying spurious signals, and excluding points forming a spur in the identified spurious signals by close proximity from limit line evaluation by comparing the points to a limit relating to the device under test. The spurious signals may be identified by identifying data points on a phase noise data curve above a noise floor determined in a preceding regression/statistical analysis. 
     The invention will be described in detail with reference to some preferred embodiments of the invention illustrated in the figures in the accompanying drawings. However, the invention is not confined to the illustrated and described embodiments alone. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Additional objects of the invention will be apparent from the following description of the preferred embodiment thereof taken in conjunction with the accompanying non-limiting drawings, in which: 
         FIG. 1  shows a block diagram of the overall system in accordance with the invention; 
         FIG. 2  reflects the receiver portion of the system in further detail; 
         FIG. 3  details a tangential correction algorithm; 
         FIG. 4  represents a loop suppression correction with polynomial fit at 100 Hz PLL loop bandwidth; 
         FIG. 5  represents a loop suppression correction with polynomial fit at 750 Hz PLL loop bandwidth; 
         FIG. 6  represents a loop suppression correction with polynomial fit at 1500 Hz PLL loop bandwidth; 
         FIG. 7   a  and  FIG. 7   b  detail a spurious/noise floor algorithm for use in the invention; 
         FIG. 7   c  outlines a proximity correction algorithm for identifying spurious in the data set for use in the invention; and 
         FIG. 8  reflects the synthesizer portion of the system in further detail. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Minimizing phase noise within a radar system provides a significant challenge to the designers of radar systems. As technology improves, phase noise specifications become more stringent in an effort to increase range detection and accuracy. Accurately measuring phase noise with repeatable results in cutting edge radar systems has proven to be even more of a challenge. Measuring phase noise down to −160 dBc or more involves measurements in the single digit nanovolt (nV) range. In order to achieve measurements down to this range, it is obvious that implementing appropriate shielding/filtering are mandatory and correction factors are necessary. Preferred embodiments of the invention will be described with reference to  FIGS. 1-8  wherein like reference numerals refer to the same or similar elements. 
     Dedicated phase noise measurement systems currently exist but are not always suited to a flight line or back shop environment. Typically, these measurement systems are not user friendly and require a high level of understanding to operate properly. As shown in the block diagram of  FIG. 1 , control of the invention is handled by a portable laptop computer  10  (which represents a general computer or other processing unit, device or system). In this embodiment, phase noise measurements are under the control of a test program set (TPS) available to be executed by the computer  10 , e.g., in a storage media of or available to the computer  10 . The TPS consists of a plurality of tests which set up the test paths and parameters and apply a set of limits. Typically, a TPS would be coded by an individual knowledgeable with the equipment and programming syntax. The TPS code is not constrained to any one language or programming environment although in a preferred embodiment, TestBASIC® is used for its ease of use and existing test executive structure which allows rapid development of a TPS. Once the TPS code has been debugged and verified, the TPS can be run by virtually anyone capable of following written and/or graphic connection prompts making it an ideal solution for flight line and back shop personnel. 
     In a preferred embodiment, the test system is controlled via an Ethernet connection  12  through a network switch  14 . Time has proven the Ethernet connection to be a robust interface with more than sufficient throughput for this application. Alternate embodiments may utilize other buses for control (such as, but not limited to: USB, MXI, FireWire, GPIB, Bluetooth or RS232 serial) or employ a plurality of buses as the application requires. In a preferred embodiment, the flight line noise tester comprises four major subsystems: a Low Noise Receiver  16 , Low Noise Synthesizer  18 , Frequency Domain Analyzer  20  (such as a spectrum analyzer) and Time Domain Analyzer  22  (such as an oscilloscope or digitizer). All of these major subsystems are connected to the network switch  14 , and thus available for control in accordance with the TPS being executed by the computer  10  Alternate embodiments may optionally include one or more DC power supplies  24  for sourcing power to the device under test (DUT)  26 . This optional feature is represented by the dotted lines in  FIG. 1 . The DUT  26  is connected to the optional DC power supplies  24 , and outputs to the Low Noise Receiver  16 . Also, in the additive mode described below, the Low Noise Synthesizer  18  provides output to the DUT  26 . 
     To facilitate transportability of the phase noise analysis system, the low noise receiver  16 , the low noise synthesizer  18 , the frequency domain analyzer  20  and the time domain analyzer  22  may be arranged in or on a transportable housing (represented by the dotted line  8 ). The optional DC supplies  24  may also be situated in or on this housing  8 . The housing  8  includes a first port to enable it to be connected to the computer  10 , the Ethernet connection  12  being a wired cable, wireless connection (WiFi) or other electrical connection feature associated with this port. The housing  8  also includes a second port to enable it to be connected to the device under test  26  for signal transfer from the device under test  26  to the low noise receiver  16 , as well as a third port to enable the output from the low noise synthesizer to be provided to the device under test  26 , i.e., in the additive mode. 
     The housing  8  also includes a fourth port to enable power from the DC power supplies  24  to be provided to the device under test  26 , i.e., to provide power thereto to facilitate the signal output for subsequent analysis. By providing the DC power supplies  24  on the housing  8 , the system can be transported to the location of the device under test  26 , coupled to the device under test  26  and computer  10  and used for phase noise analysis. This transportability is especially useful since it allows the system to be used for signal producing devices on aircraft. 
     The Low Noise Receiver  16  reflected in the block diagram of  FIG. 1  is depicted in further detail in  FIG. 2 . The RF Input from the DUT  26  enters the receiver  16  where level conditioning may be optionally applied. The signal applied at the RF Input would typically be routed to the frequency domain analyzer  20  (as shown in the depicted path) for signal identification of frequency and level. The programmable attenuator  28  allows for level reduction of higher power signals and would initially be set for maximum attenuation to prevent overload of the frequency domain analyzer  20  or other components within the receiver  16 . Gain can be added to lower level input signals via the low noise amplifier (LNA)  32  when the transfer switch  30  is switched to its opposing position as shown in the dashed inset. 
     In alternate embodiments of the invention, any transfer switch (for example, a ‘baseball’ switch) may be implemented using single pole double throw (SPDT) switches in tandem which allows for an isolation position when switches are independently controlled and set in a cross position. The input signal level can be optimally adjusted for phase noise measurement using the attenuator  28  and/or LNA  32 . This path also allows for direct spectral analysis of the phase noise on the frequency domain analyzer  20 . 
     The phase noise of an RF assembly that does not contain a signal source can be measured by the proposed invention. This is usually referred to as additive phase noise and utilizes a low phase noise signal (from the Low Noise Synthesizer  18  shown in  FIG. 1 ) fed into the DUT  26  while simultaneously feeding the Syn Input of the receiver  16  as well. 
     After level conditioning, the DUT signal passes through one of the band-dependent phase shifters  34 ,  34   a , via control of switches  35 ,  35   a . The synthesizer  18  feeds the ‘L’ port of a mixer  36  and the phase shifted DUT signal feeds the ‘R’ port of the mixer  36  with the resultant baseband (BB) signal appearing at the output of the ‘I’ port of the mixer  36 , after passing through switch  33  set in one position. In the other position of switch  33 , the signal is directed directly to the frequency domain analyzer  20  via another switch  39 . The baseband signal, containing DC+noise, is, after passing through switch  37  set in one position. filtered using a low pass filter  40  with a tunable cutoff frequency (nominally 2, 5 or 10 MHz) and routed directly or indirectly to the time domain analyzer  22 . At a minimum, the signal need only pass through a filter  41  before the time domain analyzer  22 , but alternatively may pass through other filters and/or amplifiers as described below. 
     The appropriate phase shifter is iteratively stepped over a minimum of 180° of phase shift to find the settings at which the DC component of the signal exhibits a maximum excursion as well as the minimum (as close to 0.0 V DC ). Further analysis using the frequency domain analyzer  20  at the phase shift setting corresponding to the maximum DC excursion will yield the additive amplitude modulated (AM) component of the phase noise whereas analysis at the phase shift setting corresponding to the zero DC value will yield the phase modulated (PM) component of the additive phase noise. 
     To analyze the DC+noise signal for additive phase noise, the signal is routed to the frequency domain analyzer  20  through a DC block  38  and switch  39 . The DC block  38  removes any residual DC component of the signal which could artificially skew the measurement values and/or potentially damage the frequency domain analyzer  20 . 
     The noise component must be amplified prior to measurement by the frequency domain analyzer  20 . Two low frequency LNA stages  42 ,  44  may be independently inserted using transfer switches  46 ,  48 , respectively, in a split amplification arrangement. For frequency offset measurements close to the carrier frequency (i.e. &lt;1 kHz), usually only one LNA stage would be used. Using both LNA stages at frequency offsets close in could potentially saturate the amplification resulting in artificially skewed measurement values. For carrier frequency offsets &gt;1 kHz, both LNA stages  42 ,  44  would typically be used. 
     A baseband signal containing high spurious content resulting from the AC line frequency can also saturate the LNA stage(s)  42 ,  44  and artificially skew measurement values. 
     The present invention incorporates a tunable highpass filter  50  which may be inserted prior to the LNA stages  42 ,  44  by means of a switch  52  to sufficiently reduce the line spurious content to prevent saturation of the LNA stages  42 ,  44  resulting in a measurement that accurately reflects the additive phase noise of the DUT  26 . The present invention also includes a pulse repetition frequency (PRF) filter  54  which can be inserted into the baseband path by means of a switch  56  to permit additive phase noise measurements on DUTs that operate under pulsed RF conditions. A direct BB input  58  allows baseband signals to be measured directly and may also be used for gain calibration of the LNA stages  42 ,  44 . 
     The phase noise of an RF assembly that contains a signal source can be measured by a system and method of the invention. This is usually referred to as absolute phase noise and utilizes a low phase noise signal (from the Low Noise Synthesizer  18  shown in  FIG. 1 ) fed into the Syn Input of the receiver  16  which is tuned at a nominal 10 MHz offset from the DUT carrier frequency. 
     After level conditioning, the DUT signal passes directly to the ‘R’ port of the mixer  36  along with the synthesizer on the ‘L’ port of the mixer  36 , resulting in a 10 MHz nominal difference frequency at the ‘I’ port of the mixer  36 . After passing through switch  37  set in one position, the difference frequency is bandpass filtered by filter  60 , amplified by amplifier  62  and split two ways by a signal splitter  64 ; one side of the split is feed to the ‘R’ port of the next mixer  66 , and the other side feeds into a phase lock loop (PLL) voltage controlled crystal oscillator (VCXO) module  68 . Switch  37  therefore selectively switches the output from the mixer  36  to the filter  60  or to the filter  40 . 
     The use of a phase locked crystal oscillator  70  in the PLL VCXO module  68  allows the invention to track variations in DUT carrier frequency of up to about ±22 kHz from the nominal 10 MHz VCXO frequency. In a preferred embodiment of the invention, the 10 MHz PLL VCXO module bandwidth is selectable to permit 100 Hz, 750 Hz, 1500 Hz and open loop bandwidths which allows for different DUT carrier frequency stabilities. The 10 MHz VCXO phase locks to the difference frequency of the mixer  36 , is bandpass filtered by filter  72  and fed through a phase shifter  74 . The phase shifted 10 MHz VCXO signal is amplified by amplifier  76  and filtered again, by filter  78 , and fed to the ‘L’ port of the next mixer  66 . 
     The resultant baseband (BB) signal appearing at the output of the ‘I’ port of mixer  66  contains DC+noise, is filtered using a low pass filter  80  with a tunable cutoff frequency (nominally 2, 5 or 10 MHz) and routed to the time domain analyzer  22 . The routing is dependent on the position of switches  82 ,  84 ,  86  and switches  46 ,  48 ,  52 . 
     For absolute phase noise, the baseband phase shifter  74  is iteratively stepped over a minimum of 180° of phase shift to find the settings at which the DC component of the signal exhibits a maximum excursion as well as the minimum (as close to 0.0 V DC ). Further analysis using the frequency domain analyzer  20  at the phase shift setting corresponding to the maximum DC excursion will yield the absolute amplitude modulated (AM) component of the phase noise whereas analysis at the phase shift setting corresponding to the zero DC value will yield the absolute phase modulated (PM) component of the phase noise. Analysis of the absolute DC+noise signal is the same procedure as the additive phase noise and therefore shares a common measurement path. Use of the split baseband LNA stages  42 ,  44  and bandpass filter  50  for spurious reduction is identical to that of the additive phase noise measurement. 
     A preferred embodiment of the invention also includes a downconversion stage  88  that allows signals beyond the frequency range of the phase shifters  34 ,  34   a  used for analyzing additive phase noise. In this embodiment, the DUT signal is fed into the ‘Alt DUT’ connection  90  of housing  8  and applied to the ‘R’ port of a wideband mixer  92 . One of the outputs of the low phase noise synthesizer  18  is used to feed the ‘L’ port of the mixer  92 , with the resulting downconverted signal appearing on the ‘I’ port of the mixer  92 . This signal passes through a bandpass filter  94  which is tuned to the desired high-side or low-side mix frequency. This signal is then amplified by amplifier  96  and passed out through the ‘Down Cvtrd’ connection  98  for external connection to the DUT′ connection  27  of housing  8  for additive or absolute phase noise analysis as previously described. 
     In a preferred embodiment, the invention applies additional tangential correction to a measured signal value as per the algorithm shown in  FIG. 3 . In this algorithm, the initial corrected value is set to the measured value ( 100 ) with an initial loop count of 1. The corrected value is iteratively computed ( 102 ) per the equation: 
             CV   =     MV   -     10   *       log   10     ⁡     (     1   +     10     -     ABS   ⁡     [       REF   -   CV     10     ]             )                 
until either exit condition ( 104 ,  106 ) is satisfied to provide the corrected value as the correction; if neither exit condition is satisfied the loop count is incremented ( 110 ) and the sequence repeats. One exit condition ( 104 ) is whether the difference between the current corrected value Cv i  and the immediately preceding corrected value Cv i-1  is less than 0.05, and the other exit condition ( 106 ) is whether the count is greater than 15. The algorithm in  FIG. 3  provides a software correction for tangential CV ( 108 ) on a point-by-point basis.
 
     The additional tangential correction provided by the algorithm in  FIG. 3  is exemplary only. It may not be applied in all embodiments of the invention, and alternative additional tangential correction algorithms may be alternatively or additionally applied in the invention. 
     The baseband phase lock loop allows for the tracking of time varying carrier frequencies but also has a suppression effect at low carrier offset frequencies which is more pronounced at the higher loop bandwidths. In order to achieve an accurate phase noise measurement close in to the carrier, correction for the different loop bandwidths must be applied; without correction, close in phase noise appears lower than it actually is resulting in better than expected results. 
     Proper correction for the loop bandwidth of the phase lock loop used in the absolute mode may be achieved through, for example, the use of polynomial curve fitting. The PLL suppression of the baseband signal was characterized at the specified loop bandwidths across a plurality of different types of carrier sources and compared against the open loop bandwidth setting. The aggregate data was then curve fit and a polynomial function representing the data over the applicable frequency range was generated.  FIG. 4  (100 Hz loop bandwidth),  FIG. 5  (750 Hz loop bandwidth) and  FIG. 6  (1500 Hz loop bandwidth) are representative of the data taken where the x-axis is the logarithm of the frequency offset in Hertz and the y-axis is the amplitude in dB. 
     Regarding providing a software correction for loop bandwidth suppression, the phase lock loop (PLL) suppresses the phase noise close in to the carrier. The PLL is characterized using several different types of RF sources averaged together using an independent phase noise measurement system to establish the characteristics of the PLL design. Ideally, the response should be flat at 0.0 dB amplitude (y-axis). Note that the x-axis is frequency in logarithmic terms (i.e., from  FIG. 4 , 2.0=10^2=100 Hz; 3.4=10^3.4=2512 Hz). Since the line is not flat, the line dictated by the polynomial equation is subtracted from the measured data to correct for suppression within the specified limits. 
     Phase noise measurements typically involve signals containing varying amounts of spurious content. Spurious (aka spurs) are basically defined as the random value whose variation differs from the standard error resulting from the following regression equations: 
                 ∑     n   =   1     N     ⁢           ⁢     y   i       =     bn   +     m   ⁢       ∑     n   =   1     N     ⁢           ⁢     x   i                           ∑     n   =   1     N     ⁢           ⁢       x   i     ⁢     y   i         =       b   ⁢       ∑     n   =   1     N     ⁢           ⁢     x   i         +     m   ⁢       ∑     n   =   1     N     ⁢           ⁢     x   i                 
which are re-ordered to:
 
             b   =       (       ∑           ⁢     y   i       -     m   ⁢     ∑           ⁢     x   i           )     N                 m   =       (       N   ⁢     ∑           ⁢       x   i     ⁢     y   i           -     ∑           ⁢       x   i     ⁢     ∑           ⁢     y   i             )       (       N   ⁢     ∑           ⁢     x   i   2         -       (     ∑           ⁢     x   i       )     2       )             
substituting the following:
 
     
       
         
           
             
               S 
               xx 
             
             = 
             
               
                 N 
                 ⁢ 
                 
                   ∑ 
                   
                       
                   
                   ⁢ 
                   
                     x 
                     i 
                     2 
                   
                 
               
               - 
               
                 
                   ( 
                   
                     ∑ 
                     
                         
                     
                     ⁢ 
                     
                       x 
                       i 
                     
                   
                   ) 
                 
                 2 
               
             
           
         
       
     
               S   yy     =       N   ⁢     ∑           ⁢     y   i   2         -       (     ∑           ⁢     y   i       )     2                     S   xy     =       N   ⁢     ∑           ⁢       x   i     ⁢     y   i           -     ∑           ⁢       x   i     ⁢     ∑           ⁢     y   i                   
results in:
 
             m   =       S   xy       S   xx             
and forms the equation:
 
 y   i   =b+mx+ε   i  
 
where ε i  is a normally distributed random variable with zero mean and common variance σ 2 . The standard error is then defined as:
 
     
       
         
           
             
               S 
               e 
               2 
             
             = 
             
               
                 1 
                 
                   ( 
                   
                     N 
                     - 
                     2 
                   
                   ) 
                 
               
               ⁢ 
               
                 ∑ 
                 
                     
                 
                 ⁢ 
                 
                   
                     ( 
                     
                       
                         y 
                         i 
                       
                       - 
                       
                         ( 
                         
                           b 
                           + 
                           
                             mx 
                             i 
                           
                         
                         ) 
                       
                     
                     ) 
                   
                   2 
                 
               
             
           
         
       
     
     In a preferred embodiment, raw phase noise measurement data is analyzed for spurious content using regression/statistical algorithms through the use of the above equations applied in the algorithm shown in  FIGS. 7   a  and  7   b . Three user variables (USER1, USER2, USER3; each having a nominal value of 1) allow for adjustment under varying conditions of spurious content. An explanation of the manner in which the algorithm is executed is easily gleaned from the content of  FIGS. 7   a  and  7   b , which steps are based on the above equations. This algorithm is not limiting and other comparable or equivalent algorithms that achieve the same function of identifying spurious signals may be applied in the invention. 
     Once the spurious signals have been identified, all points forming a spur by close proximity are excluded from limit line evaluation.  FIG. 7   c  outlines the algorithm as to determining which points are excluded from evaluation. The excluded points are essentially the data points on the phase noise data curve identified as spurious which are above the noise floor (NF_Line) determined in the prior regression/statistical analysis. Points falling outside of the identified spurs are subject to the user limit lines established for the DUT  26 . An explanation of the manner in which the algorithm in  FIG. 7   c  is executed is easily gleaned from the content of  FIG. 7   c , which steps are based on the above equations. This algorithm is not limiting and other comparable or equivalent algorithms that achieve the same function of excluding points forming a spur by close proximity from limit line evaluation may be applied in the invention. In order to measure low phase noise, a stable low phase noise source is required on the ‘Syn’ input to the receiver  16 . The invention includes a programmable synthesizer module (embodied as or within the low noise synthesizer  18 ) capable of producing a broad range of frequencies necessary for measuring additive and absolute phase noise. In order to achieve the lowest phase noise possible, a variety of different techniques including surface acoustic wave (SAW) oscillators, oven controlled crystal oscillators (OCXO) and a direct digital synthesizer (DDS) are employed. OCXO oscillators inherently have good phase noise at carrier frequency offsets of less than 100 kHz whereas SAW oscillators have good phase noise at carrier frequency offsets greater than 100 kHz. Since OCXO and SAW oscillators have fixed frequency outputs, the addition of a DDS element results in a low phase noise source with a programmable frequency when it is mixed with the OCXO or SAW oscillators. 
     Referring now to  FIG. 8 , the invention employs a plurality of surface acoustic wave (SAW) oscillators  112 ,  114  to generate signals with low phase noise at carrier frequency offsets greater than about 100 kHz. The first SAW oscillator  112 , having a nominal fundamental frequency of about 960 MHz is amplified by amplifier  116  and low pass filtered by filter  118  and can directly feed the ‘L’ port of a mixer  120  via a switch  119  or the ‘L’ port of a mixer  122  depending on the setting of switch  119 , as well as the settings of switches  134 ,  136 . This signal may also be switched through a transfer switch  124  where a frequency multiplier or harmonic generator  126  multiplies the fundamental frequency in integer multiples (i.e. ×2, ×3, etc.) using step recovery diode (SRD), overdriven amplifier or similar means. It is assumed the practical upper frequency limit of any frequency multipliers discussed herein is nominally 20 GHz mainly due to cost constraints but this is not intended to be a limiting factor of the invention. After frequency multiplication, the signal is filtered using a tunable bandpass filter  128  to select the desired harmonic necessary and amplified by amplifier  130  before being applied to either mixer  120 ,  122 . 
     In the embodiment shown in  FIG. 8 , the second SAW oscillator  114  having a nominal fundamental frequency of about 1090 MHz can directly feed the ‘I’ port of the mixer  120  through a switch  132  or the ‘L’ port of the mixer  122  through switches  132 ,  134 ,  136 . The signal from the second SAW oscillator  114  can be subjected to the same processing as the signal from the first SAW oscillator  112 , e.g., frequency multiplied and processed by components  116   a ,  118   a ,  124   a ,  126   a ,  128   a ,  130   a  comparable to respective components  116 ,  118 ,  124 ,  126 ,  128  and  130 , and mixed with the first SAW oscillator chain with the resultant frequency appearing at the ‘R’ port of the mixer  120 . In this arrangement, the resultant signal from mixer  120  is filtered using a tunable bandpass filter  138  to select the desired mixer product and amplified by amplifier  140  before being applied to mixer  122 . 
     The invention also employs a plurality of oven controlled crystal oscillators (OCXO)  142 ,  144 ,  146  to generate signals with low phase noise at carrier frequency offsets less than about 100 kHz. The first OCXO  142 , having a nominal fundamental frequency of about 100 MHz can directly feed the ‘L’ port of the mixer  148  through switches  151 ,  152  or the ‘L’ port of the mixer  122  through switches  151 ,  152 ,  156 ,  136 . The OCXO  142  can be frequency multiplied in a fashion similar to the SAW oscillators so that higher frequencies can be applied to mixer  148  or mixer  122 . This may be implemented by the transfer switch  151 , leading to/from a frequency multiplier generator  158 , a tunable bandpass filter  160  and an amplifier  162 . The ‘R’ port of mixer  148  may be bandpass filtered by tunable filter  149  to select the desired mixer product, and then amplified by amplifier  150  before being applied to mixer  122  via switches  154 ,  156  and  136 . 
     The second OCXO  144  in the embodiment shown in  FIG. 8  is intended to operate a little differently. This OCXO  144 , having a nominal fundamental frequency of about 120 MHz, provides a signal that passes through a power splitter  164 ; one side of the split is frequency multiplied by frequency multiplier  166  by default, filtered using a tunable bandpass filter  168  and amplified by amplifier  170  to select the desired harmonic necessary. In a preferred embodiment, this multiplication factor will be 8 resulting in a 960 MHz carrier frequency. This signal can directly feed the ‘I’ port of mixer  148  through switch  154  or the ‘L’ port of mixer  122  through switches  154 ,  156 ,  136 . The multiplied carrier passes through an RF coupler  171  which is used to source the reference frequency to the DDS  172 . 
     The DDS  172  is capable of operation from DC to nominally 40% of the reference frequency. The upper frequency limit is dictated by Nyquist sampling theory which, in this case, would be about 50% of the reference frequency. In a preferred embodiment, this upper frequency limit would be somewhere around about 400 MHz. The output produced by the DDS  172  is then amplified by amplifier  174  and filtered using a tunable bandpass filter  176  to reduce harmonic content. This signal can directly feed the ‘I’ port of mixer  178  via switch  180  or amplified by amplifier  182  and routed to the ‘I’ port of mixer  122 . 
     The second side of the OCXO splitter  164  can directly feed the ‘L’ port of mixer  178  via switches  184 ,  186  or switched by switch  186  into another multiplier chain  188 . The multiplier chain  188  includes a frequency multiplier  190 , a tunable bandpass filter  192  and an amplifier  194 . 
     In the embodiment shown in  FIG. 8 , the third OCXO  146 , having a nominal frequency of about 160 MHz, may provide an output signal that can be routed to the multiplier chain  188  in lieu of the signal from the second OCXO  144 . The resultant frequency appearing at the ‘R’ port of the mixer  178  is filtered using a tunable bandpass filter  196  to select the desired mixer product and amplified by amplifier  182  before being applied to mixer  122 . Alternate embodiments may introduce interim connection points at any of SAW, OCXO and DDS stages using switches, splitters, couplers, etc. as user requirements dictate. 
     The signal produced at the ‘R’ port of mixer  122  is capable of being programmed to a broad range of frequencies through selection of the various multiplier factors, bandpass filter frequencies and the DDS. Phase noise performance can be tailored for low frequency offset phase noise performance by primarily using the OCXO-based sources ( 112 ,  114 ) or for higher frequency offset phase noise performance using the SAW oscillator-based sources ( 142 ,  144 ,  146 ). This signal is bandpass filtered by tunable filter  198  to select the desired mixer product and amplified by amplifier  200  before being applied to a splitter  202 . The first leg from the splitter  202  is routed to a second splitter  204  which is capable of being routed to two outputs simultaneously via switches  206 ,  208 . The second leg from the splitter  202  passes through a multiplier chain  210 , including a frequency multiplier  212 , tunable bandpass filter  214  and amplifier  216 , and then to another splitter  218  which is also capable of being routed to same two outputs via switches  206 ,  208 . These two outputs may consist of the fundamental output of mixer  122 , a frequency multiplied version of the mixer output or combination of both. The following phase noise performance is achievable: 
     
       
         
               
               
               
             
           
               
                   
               
               
                 Offset 
                 L-Band 
                 S-Band 
               
               
                   
               
             
             
               
                 100 Hz   
                 −109 dBc/Hz 
                  −99 dBc/Hz 
               
               
                  1 kHz 
                 −133 dBc/Hz 
                 −123 dBc/Hz 
               
               
                  10 kHz  
                 −142 dBc/Hz 
                 −132 dBc/Hz 
               
               
                 100 kHz 
                 −151 dBc/Hz 
                 −141 dBc/Hz 
               
               
                 500 kHz 
                 −159 dBc/Hz 
                 −149 dBc/Hz 
               
               
                  1 MHz 
                 −159 dBc/Hz 
                 −149 dBc/Hz 
               
               
                  2 MHz 
                 −159 dBc/Hz 
                 −149 dBc/Hz 
               
               
                   
               
             
          
         
       
     
     The flight line tester included in housing  8  is intended to test military radar systems or subassemblies thereof, i.e., which would be the device under test  26 . In a broader sense, the tester is capable of testing RF sources (absolute phase noise) or other RF assemblies (additive phase noise). The device under test could be any of these, e.g., an RF source of an RF assembly. 
     Having thus described a few particular embodiments of the invention, various alterations, modifications, and improvements will readily occur to those skilled in the art. Such alterations, modifications and improvements as are made obvious by this disclosure are intended to be part of this description though not expressly stated herein, and are intended to be within the spirit and scope of the invention. Accordingly, the foregoing description is by way of example only, and is not limiting. The invention is limited only as defined in the claims and equivalents thereto.