Abstract:
An RF power amplifier with variable bias current is disclosed. The RF amplifier includes a peak detector that detects the peak level of the amplifier input signal. The peak detector generates an output signal in response to the peak level of the amplifier input signal. A bias voltage level setting circuit coupled to the peak detector receives the peak detector output signal and generates a bias voltage in response to the peak detector output signal. An amplifier circuit coupled to the bias voltage level setting circuit receives the bias voltage and the amplifier input signal, and generates an output signal in response to the bias voltage and the amplifier input signal. The disclosed RF amplifier allows amplification of RF signals with high linearity and high efficiency at varying power levels, and extends the maximum power capability of the amplifier.

Description:
TECHNICAL FIELD OF THE INVENTION 
     The present invention relates to RF signal processing, and in particular to an RF power amplifier with variable bias current. 
     BACKGROUND OF THE INVENTION 
     In cellular telephones and other communication devices, radio frequency (RF) power amplifiers are typically used to amplify RF signals prior to transmission. These RF power amplifiers typically generate an output power in the range of 50 mW to 3 watts. In such devices, linear amplification is desired to prevent signal distortion. Efficiency is also a consideration, especially for mobile devices such as cellular telephones, due to the limited quantity of energy stored in the accompanying battery. 
     Efficiency and linearity are often competing considerations. When high efficiency is important, a low amplifier transistor bias current is chosen, thereby increasing battery life and talk time. This generally results in acceptable distortion at low to moderate power levels, but creates unacceptable distortion at high power levels. When high linearity is important, a larger transistor bias current is chosen, reducing distortion to an acceptable level even at high power levels. The high bias current may also be required to obtain the maximum output power from the amplifier output transistor. However, the high bias current reduces battery life and talk time, particularly at low power levels. 
     SUMMARY OF THE INVENTION 
     Therefore, a need has arisen for an RF power amplifier that addresses the disadvantages and deficiencies of the prior art. In particular, a need has arisen for an RF power amplifier with high efficiency and high linearity at varying power levels without the need for external adjustment. 
     Accordingly, an RF power amplifier with variable bias current is disclosed. In one embodiment, the RF amplifier includes a peak detector that detects the peak level of the amplifier input signal. The peak detector generates an output signal in response to the peak level of the amplifier input signal. A bias voltage level setting circuit coupled to the peak detector receives the peak detector output signal and generates a bias voltage in response to the peak detector output signal. An amplifier circuit coupled to the bias voltage level setting circuit receives the bias voltage and the amplifier input signal, and generates an output signal in response to the bias voltage and the amplifier input signal. 
     A technical advantage of the present invention is that amplification of RF signals may be obtained with high linearity at varying power levels. Another technical advantage of the present invention is that high linearity may be attained while maintaining low bias current and high efficiency at varying power levels. Yet another technical advantage of the present invention is that high maximum output power may be obtained with high efficiency. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     For a more complete understanding of the present invention and for further features and advantages, reference is now made to the following description taken in conjunction with the accompanying drawings, in which: 
     FIG. 1 is a simplified perspective view of a mobile telephone constructed in accordance with the present invention; 
     FIG. 2 is a schematic diagram of an RF power amplifier constructed in accordance with one embodiment of the present invention; 
     FIG. 3 is a graph illustrating waveforms in the RF power amplifier; and 
     FIG. 4 is a schematic diagram of an alternative RF power amplifier constructed in accordance with one embodiment of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     The preferred embodiments of the present invention and their advantages are best understood by referring to FIGS. 1 through 4 of the drawings. Like numerals are used for like and corresponding parts of the various drawings. 
     Referring to FIG. 1, a mobile telephone  1  constructed in accordance with the present invention is shown. Mobile telephone  1  has a microphone  2 , a speaker  3 , a keypad  4 , a display screen  5 , and a radio frequency (RF) antenna  6  for sending and receiving signals from a station such as a cell tower (not shown). Mobile telephone  1  also has internal circuitry  7  powered by a battery  8 . Mobile telephone  1  may be compliant with a signal frequency and modulation standard such as AMPS, PCS, GSM, CDMA, TDMA, DCS 1800 or some other telecommunications standard. 
     Internal circuitry  7  is coupled to speaker  3  and microphone  2  for communicating with a user. Internal circuitry  7  is also coupled to keypad  4  to receive information regarding keypad entries made by the user. Internal circuitry  7  is also coupled to RF antenna  6  to send and receive identification signals, voice signals, keypad entries and other information to and from the station. 
     Internal circuitry  7  communicates with the station via RF signals transmitted through the atmosphere. To generate RF signals, internal circuitry  7  includes one or more RF power amplifiers (not shown in FIG. 1) capable of amplifying RF signals. 
     Referring to FIG. 2, an RF power amplifier  10  for use in internal circuitry  7  of mobile telephone  1  or in other RF devices is shown. RF power amplifier  10  has a variable bias current, as will be described more fully below. RF power amplifier  10  is preferably implemented in a GaAs MESFET technology, but may also be implemented in a Si MOSFET, Si bipolar, GaAs HBT or some other technology. 
     An input signal V in  is provided to RF power amplifier  10  by an input signal source  12  with a source impedance R s  represented by resistor  14 . Input signal source  12  is coupled to a DC blocking capacitor  16 . Capacitor  16  may also form an input impedance matching network with an inductor  18 . Thus, capacitor  16  and inductor  18  together may provide an input impedance which preferably matches the source impedance R s  of input signal source  12  to the input impedance of an amplifier transistor  32 , which will be described more fully below. This match may be for maximum gain, maximum output power, best linearity, or some combination of these parameters. 
     A diode  20  has a cathode connected to one terminal of capacitor  16  and an anode connected to one terminal of a holding capacitor  22 . The other terminal of holding capacitor  22  is grounded. Together, diode  20  and holding capacitor  22  form a negative peak detector  23 . The most negative voltage seen at the cathode of diode  20  (plus a diode drop) is held by holding capacitor  22  at node  24 , between holding capacitor  22  and diode  20 . 
     The voltage at node  24  (V 24 ) is provided to a low-pass feedback amplifier  31  formed by control amplifier  26 , resistor  28  and capacitor  30 . Resistor  28  has a resistance R 1  and is connected between node  24  and the negative input terminal of control amplifier  26 . Capacitor  30  is connected between the output terminal and the negative input terminal of control amplifier  26 . Control amplifier  26  may be an operational amplifier or some other amplifier with a gain preferably no less than 10. A positive reference voltage V ref  is provided to the positive input terminal of control amplifier  26  by a voltage source  27 . 
     In this configuration, control amplifier  26 , together with resistor  28  and capacitor  30 , acts as an integrator that integrates (V 24 −V ref ) over time. Thus, if V 24  is less than V ref , the output of control amplifier  26  will increase. Similarly, if V 24  is greater than V ref , the output of control amplifier  26  will decrease. For reasons which will become apparent, the response time of low-pass feedback amplifier  31 , as determined by the relative sizes of capacitors  22  and  30  and resistor  28 , is preferably at a rate that is slower than the modulating baseband signal. 
     The output of control amplifier  26  provides a bias voltage to the gate of a transistor  32 . Thus, AC signals received from signal source  12  via capacitor  16  are added to the quasi-constant bias voltage generated by control amplifier  26 . The response time of low-pass feedback amplifier  31  is preferably at a rate that is slower than the modulating baseband signal so that the bias voltage created by low-pass feedback amplifier  31  does not unduly distort the envelope of the RF signal provided to transistor  32 . Inductor  18 , connected between the gate of transistor  32  and the output of control amplifier  26 , and an RF bypass capacitor  34 , connected between the output of control amplifier  26  and ground, prevent the RF signal from signal source  12  from affecting the function of low-pass feedback amplifier  31 . 
     Transistor  32  can be modeled as a transconductance amplifier with a threshold voltage V t . Thus, when the gate voltage (V gs ) is higher than V t , transistor  32  conducts a drain current (I d ) proportional to a function of V gs −V t , specifically (V gs −V t ) q , where 1&lt;q&lt;3. When V gs &lt;V t , I d ≈0. The drain current I d  is converted to an output voltage V out  by a load resistor  36 . An inductor  38  acts as an RF choke. An inductor  40  and capacitor  42  form an output impedance matching network. Capacitor  44  acts as a DC blocking capacitor. 
     The bias voltage supplied to the gate of transistor  32  determines the quiescent drain current I d  conducted by transistor  32  and affects the average transistor current under conditions of RF drive. In previous RF power amplifier circuits, a constant bias voltage, typically greater than the threshold voltage V t  of the amplifier transistor, was provided to the amplifier transistor. The constant bias voltage was typically preselected by the circuit designer as a compromise between efficiency, linearity and maximum output power. Thus, if the biasing of the amplifier transistor was such that the minimum V gs  was approximately equal to V t , then drain current I d  would be cut off at each downswing of V gs . Such an amplifier is referred to as a Class B amplifier. In this situation, some distortion is usually introduced into the output voltage V out . In CDMA and TDMA environments, this distortion may result in spectral regrowth, causing loss of information and spillover into adjacent channels. Conversely, if a high bias voltage was selected to prevent clipping of the drain current waveform, linearity and/or maximum output power may be improved but a greater average drain current would result, decreasing battery life and talk time. 
     In RF power amplifier  10 , a variable bias voltage is produced by low-pass feedback amplifier  31 . The operation of RF power amplifier  10  is illustrated in FIG. 3, in which a waveform  50  of input voltage V in  and a corresponding waveform  52  of gate voltage V gs  are shown. In FIG. 3, it is assumed for purposes of illustration that an optional resistor  46  in RF power amplifier  10  has an infinite resistance, or in other words that resistor  46  is absent from RF power amplifier  10 . The effect of resistor  46  will be described more fully below. 
     As shown in FIG. 3, input voltage V in  initially has a sinusoidal waveform with a first amplitude. A bias voltage V bias1  is produced by low-pass feedback amplifier  31  so that V gs  does not fall below V ref . At time t 0 , the amplitude of V in  increases. This increase in amplitude is also seen in V gs . At time t 1 , peak detector  23  detects a minimum V gs  below V ref . This imbalance between the minimum V gs  and V ref  causes low-pass feedback amplifier  31  to increase the bias voltage applied to the gate of transistor  32 . Over time, the bias voltage is increased until, at time t 2 , a bias voltage V bias2  is reached such that the minimum V gs  detected by peak detector  23  is equal to V ref . This bias level V bias2  is maintained so long as the amplitude of V in  remains constant. If the amplitude of V in  later increases or decreases, low-pass feedback amplifier  31  will increase or decrease the bias voltage accordingly. 
     Reference voltage V ref  is preferably chosen to be greater than or approximately equal to the threshold voltage V t  of transistor  32 . Thus, at any constant power level, amplifier  10  acts as a Class A amplifier, with no cutoff of drain current I d . Class B operation occurs only on a temporary basis immediately following an increase in power level, before a new bias voltage level has been attained by low-pass feedback amplifier  31 . Low-pass feedback amplifier  31  preferably has a response time that is slower than the modulating baseband signal, so that the adjustment of bias voltage level by low-pass feedback amplifier  31  does not unduly distort the envelope of the gate voltage waveform. 
     Referring again to FIG. 2, an optional resistor  46  with a resistance R 2  is connected between the negative input of control amplifier  26  and the gate of transistor  32 . In the foregoing description of the operation of RF power amplifier  10 , it was assumed that resistor  46  was not present, i.e. that R 2 =∞. With resistor  46  present, the behavior of low-pass feedback amplifier  31  under constant input power conditions can be approximated by the following equation: 
     
       
           V   min   =V   ref   −V   p   R   1 /( R   1   +R   2 )  (1) 
       
     
     In equation (1), V min  is the minimum gate voltage V gs  detected by peak detector  23  and V p  is the peak amplitude of the input voltage V in . 
     Thus, for the ideal case previously discussed, in which R 2 =∞, V min =V ref . In other words, the steady-state minimum V gs  will be equal to V ref , as illustrated in FIG.  3 . If R 1 =∞, so that low-pass feedback amplifier  31  is isolated from peak detector  23 , then a constant bias voltage equal to V ref  is produced, so that V min =V ref −V p . This is similar to a typical Class B or other constant-bias amplifier. 
     If both R 1  and R 2  are finite, then, according to equation (1), the bias voltage produced by low-pass feedback amplifier  31  produces a bias condition somewhere between Class A operation (with no drain current cutoff) and Class B operation. This mode of operation, referred to a “Class A/B” operation, typically results in improved linearity or efficiency compared to Class A or Class B operation. Thus, some drain current cutoff may occur at high input power, but less drain current is used, resulting in longer battery life and talk time. The circuit designer may therefore select R 1  and R 2  to achieve more optimum performance or a compromise between efficiency and linearity, with the resulting compromise giving acceptable efficiency and linearity over a greater range of input power than could be attained with a constant-bias amplifier. 
     In RF power amplifier  10 , when diode  20  conducts a current, there is a temperature-dependent voltage drop across the diode. Ideally, peak detector  23  detects the minimum V gs  seen at the gate of transistor  32 . However, the voltage drop across diode  20  creates an offset between V 24  and the minimum V gs . Furthermore, because the voltage drop across diode  20  is dependent on diode temperature, the offset between V 24  and the minimum V gs  is dependent on the ambient temperature experienced by RF power amplifier  10 . This temperature effect on the performance of RF power amplifier  10  is undesirable. 
     Thus, referring to FIG. 4, an alternative RF power amplifier  50  with variable bias current is shown. RF power amplifier  50  operates in a manner similar to RF power amplifier  10 , but reduces or eliminates the aforementioned disadvantages of RF power amplifier  10 . Like RF power amplifier  10 , RF power amplifier  50  receives an input signal V in  with a source impedance R s  from an input signal source  52  via a DC blocking capacitor  54 . Capacitor  54  also forms an input impedance matching network with an inductor  56 . Thus, capacitor  54  and inductor  56  together provide an input impedance which preferably matches the source impedance R s  of input signal source  52  to the input impedance of amplifying transistor  80  in a desired way. 
     A diode  58  and holding capacitor  60  form a negative peak detector  64 . Unlike peak detector  23  of RF power amplifier  10 , peak detector  64  holds a closer approximation of the most negative voltage seen at the cathode of diode  58  with a substantially reduced diode drop. This is because current sources  66  and  68  each conduct a small current so as to set diode  58  at the edge of conduction. Thus, the most negative voltage seen at the cathode of diode  58  is held by holding capacitor  60  at node  70  between holding capacitor  60  and diode  58 . 
     The voltage at node  70  (V 70 ) is provided to a low-pass feedback amplifier  71  formed by control amplifier  72 , resistor  74  and capacitor  76 . A positive reference voltage V 1  is provided to the positive input terminal of control amplifier  72  by a voltage source  78 . Low-pass feedback amplifier  71  integrates (V 70 −V 1 ) over time. Thus, if V 70  is less than V 1 , the output of control amplifier  72  will increase. Similarly, if V 70  is greater than V 1 , the output of control amplifier  72  will decrease. 
     A diode  77  connected between voltage source  78  and the positive input terminal of control amplifier  72  is set at the edge of conduction by a current source  79 . Diode  77  preferably has temperature-dependent characteristics similar to those of diode  58 . Thus, because each input of control amplifier  72  is similarly affected by temperature variations, the performance of low-pass feedback amplifier  71  is largely temperature-independent. 
     Similarly, a diode  96  is connected between resistor  94  and the negative input of control amplifier  72 . Diode  96  is set at the edge of conduction by current sources  98  and  99 . Diode  96 , which preferably has temperature characteristics similar to those of diode  58 , compensates for the temperature dependence of diode  58 . Diode  96 , together with diode  77 , make equation (1) a better approximation of the behavior of low-pass feedback amplifier  71 . 
     The output of control amplifier  72  provides a bias voltage to the gate of a transistor  80  and is isolated at RF frequencies. Thus, AC signals received from signal source  52  via capacitor  54  are added to the quasi-constant bias voltage produced by control amplifier  72 . The response time of low-pass feedback amplifier  71  is preferably at a rate that is slower than the modulating baseband signal so that the bias voltage created by low-pass feedback amplifier  71  does not unduly distort the envelope of the RF signal provided to transistor  80 . Inductor  56 , connected between the gate of transistor  80  and the output of control amplifier  72 , and an RF bypass capacitor  82 , connected between the output of control amplifier  72  and ground, prevent the RF signal from signal source  52  from affecting the function of low-pass feedback amplifier  71 . 
     Transistor  80  can be modeled as a transconductance amplifier with a threshold voltage V t  and an appropriate output impedance. Thus, when the gate voltage (V gs ) is higher than V t , transistor  80  conducts a drain current (I d ) proportional to a function of V gs −V t . When V gs &lt;V t , I d =0. The drain current I d  is converted to an output voltage V out  by a load resistor  84 . An inductor  86  acts as an RF choke. An induct or  88  and capacitor  90  form an output impedance matching network. Capacitor  92  acts as a DC blocking capacitor. 
     One preferred mode of operations involves setting reference voltage V ref  to be greater than or approximately equal to the threshold voltage V t  of transistor  80 . Thus, at any constant power level, amplifier  50  acts approximately as a Class A amplifier, with no cutoff of drain current I d . As previously discussed with respect to FIG. 3, Class B operation occurs only on a temporary basis immediately following an increase in power level, before a new bias voltage level has been attained by low-pass feedback amplifier  71 . However, low-pass feedback amplifier  71  preferably has a response time that is slower than the modulating baseband signal, so that the adjustment of bias voltage level by low-pass feedback amplifier  71  does not unduly distort the waveform of the gate voltage V gs . 
     An optional resistor  94  with a resistance R 2  is coupled between the negative input of control amplifier  72  and the gate of transistor  80 . With resistor  94  present, the behavior of low-pass feedback amplifier  71  under constant input power conditions can be approximated by equation (1) as previously discussed. 
     If both R 1  and R 2  are finite, then, according to equation (1), the bias voltage produced by low-pass feedback amplifier  71  results in Class A/B operation. The exact choice of R 1  and R 2 , along with the choice of output impedance, is generally made to achieve a satisfactory combination of efficiency, linearity and maximum output power. 
     Although the present invention and its advantages have been described in detail, it should be understood that various changes, substitutions, and alterations can be made therein without departing from the spirit and scope of the invention as defined by the appended claims.