Abstract:
Apparatus and methods for enhancement of performance in polar modulators are described. In one implementation, an FM signal component is divided into a coarse FM signal and a residue FM signal to be applied to a VCO for generation of a modulation signal. The coarse FM signal may be proportional to a threshold value when the FM signal exceeds the threshold, while the residue FM signal may be proportional to the difference between the FM signal and coarse FM signal.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application claims priority under 35 U.S.C. §119(e) to U.S. Provisional Patent Application Ser. No. 60/992,294, entitled POLAR MODULATION WITH FM SPLITTING, filed on Dec. 4, 2007. This application is also related to U.S. Utility Pat. No. 6,985,703, entitled DIRECT SYNTHESIS TRANSMITTER, issued Jan. 10, 2006, to U.S. Utility Pat. No. 6,774,440 entitled SYSTEM FOR HIGHLY LINEAR PHASE MODULATION, issued Aug. 10, 2004, to U.S. Utility Pat. No. 7,764,139, entitled POLAR MODULATION WITH EXTENDED AM, issued Jul. 27, 2010, to U.S. Utility patent application Ser. No. 12/251,342, entitled APPARATUS AND METHODS FOR FM PULSE SHAPING, filed Oct. 14, 2008, to U.S. Utility patent application Ser. No. 12/249,725 entitled APPARATUS AND METHODS POLAR MODULATION WITH IQ ZEROING, filed Oct. 10, 2008, and to U.S. Utility Pat. No. 7,675,379, entitled LINEAR WIDEBAND PHASE MODULATION SYSTEM, filed Mar. 9, 2010. The content of each of these applications is hereby incorporated by reference herein in its entirety for all purposes. 
     FIELD OF THE INVENTION 
     The present invention relates generally to radio transmitters using polar modulation. More particularly but not exclusively, the present invention relates to apparatus and methods for reducing peak analog deviation to improve performance and ease voltage controlled oscillator (VCO) design requirements. 
     BACKGROUND 
     Radio transmitters are used to generate the modulated signals required for wireless communications using modulation techniques such as QPSK, 8-PSK, 16-QAM, 64-QAM, and OFDM to vary the amplitude, phase, and/or frequency of the transmitter&#39;s RF carrier. 
     The modulated signal represents and conveys the message data consisting of in phase (I) and quadrature (Q) data streams. In practice, these data streams pass through digital filters that shape the resulting pulses and ultimately define the spectrum of the modulated transmit signal. A polar transmitter translates these I and Q data streams to equivalent amplitude (AM) and phase (PM) modulation signals. This allows these signals to be applied at more advantageous points in the transmitter increasing its efficiency. 
     The PM signal is applied to the RF carrier at a phase-locked loop (PLL). In practice, this is actually accomplished using the equivalent frequency modulation (FM) signal, which is easily found by differentiating the PM signal. Unfortunately, the differentiation process widens the bandwidth of the FM signal and also generates impulses. This is due to the fact that the phase jumps by as much as π whenever the transmit signal passes through or near the origin of the complex plane as shown in  FIG. 1 . The resulting FM impulses (that occur after differentiating the phase jumps), although infrequent, can be as strong as one-half of the data rate. 
     The FM signal&#39;s impulses and wide bandwidth present daunting challenges to the design of the polar transmitter. Any distortion of the FM signal alters the spectrum of the VCO output, elevates the noise floor around the transmit signal, and rotates the complex signal pattern. Practical circuits invariably reduce the bandwidth of the FM signal and degrade performance. More importantly, the VCO and PLL limit the peak FM deviation and corrupt the transmit output spectrum. It would therefore be advantageous to reduce the peak FM deviation as well as the bandwidth of the FM signal. 
     SUMMARY 
     In one or more embodiments of the present invention, apparatus and methods for dividing an FM signal in a polar modulation transmitter into a coarse digital term and a fine analog residue to better realize the large frequency deviations required of the VCO and phase/frequency modulator are described. 
     In one aspect, the present invention relates to a method for processing an FM signal component of a polar modulation signal, comprising comparing the FM signal to a threshold value, and, responsive to said comparing, generating a residue FM signal, wherein said residue FM signal is proportional to the difference between said FM signal and said threshold value when said FM signal exceeds said threshold value and proportional to the FM signal when said FM signal is less than said threshold value. 
     In another aspect, the present invention relates to a method for processing an FM signal component of a polar modulation signal, comprising comparing the FM signal to one of a plurality of threshold values, and, responsive to said comparing, generating a residue FM signal, wherein said residue FM signal is proportional to the difference between said FM signal and one or more of said plurality of threshold values when said FM signal exceeds one or more of said threshold values and proportional to the FM signal when said FM signal is less than said one or more threshold values. 
     In yet another aspect the present invention relates to an Apparatus for processing an FM signal component of a modulation signal in a polar modulator, comprising a detection circuit configured to determine when the FM signal exceeds a threshold value, a coarse FM signal generation circuit configured to generate, responsive to said determining, a coarse FM signal representative of said threshold value when said FM signal exceeds said predetermined threshold, and a residue FM signal generation circuit configured to generate, responsive to said determining, a residue FM signal having a value proportionate to the difference between said FM signal and said threshold value when said FM signal exceeds said threshold value. 
     In yet another aspect, the present invention relates to an Apparatus for processing an FM signal component of a modulation signal in a polar modulator, comprising a detection circuit configured to determine when the FM signal exceeds a threshold value, a coarse signal generation circuit configured to generate, responsive to said determining, one or more coarse FM signals representative of a plurality of threshold values when said FM signal exceeds one or more predetermined thresholds, and a residue signal generation circuit configured to generate, responsive to said determining, a residue FM signal having a value proportionate to the difference between said FM signal and one of said plurality of threshold values when said FM signal exceeds said plurality of threshold values. 
     Additional aspects of the present invention are described below with respect to the appended drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The following is a brief description of the drawings wherein: 
         FIG. 1  shows the complex signal trajectory for a WCDMA transmit signal; 
         FIG. 2  shows a diagram of simple polar transmitter; 
         FIG. 3(   a ) shows a diagram of a fractional-N PLL with a A modulator; 
         FIG. 3(   b ) shows noise contributions of the A modulator of  FIG. 3(   a ); 
         FIG. 4(   a ) shows a phase/frequency modulator; 
         FIG. 4(   b ) shows the responses associated with each of the modulation paths of the modulator of  FIG. 4(   a ); 
         FIG. 5(   a ) shows a dual port VCO; 
         FIG. 5(   b ) shows the response at the modulation port for the VCO of  FIG. 5(   a ); 
         FIG. 6  shows the FM signal component for a typical WCDMA transmit signal; 
         FIG. 7(   a ) shows an embodiment of FM splitting in accordance with aspects of the present invention; 
         FIG. 7(   b ) shows an embodiment of a modified VCO in accordance with aspects of the present invention; 
         FIG. 7(   c ) shows resulting digital and analog signals for an embodiment of the present invention. 
     
    
    
     DETAILED DESCRIPTION 
     A simple diagram of a polar transmitter is shown in  FIG. 2 . It synthesizes the transmit signal using direct phase modulation at the synthesizer and amplitude modulation at the variable gain amplifier (VGA) or power amplifier (PA). 
       FIG. 3(   a ) shows a fractional-N phase-locked loop (PLL) used to synthesize the radio frequency (RF) carrier signal. The PLL forms a feedback system that consists of a voltage-controlled oscillator (VCO), N counter, phase/frequency detector (P/FD), charge pump (CP), and integration filter. 
     The PLL uses negative feedback to force the phase of the feedback signal to track the phase of the reference signal. As a result, the VCO oscillates at a frequency given by
 
 f   VCO   =f   REF ( N+n )
 
where n represents the fractional value and N equals the integer value.
 
     The fractional-N phase-locked loop resolves fine frequency steps by modulating the value of Δn so that its average value satisfies 
             n   =         Δ   ⁢           ⁢   f       f   REF       =       1   M     ⁢       ∑     i   =   1     M     ⁢     Δ   ⁢           ⁢     n   i                   
The ΔΣ modulator forms a sequence of Δn values with these important properties: 1) it responds to the input n quickly, 2) it possesses a resolution that improves with the number of samples, and 3) it concentrates quantization noise at high frequencies, near one-half the clock frequency.
 
     The quantization noise can be attributed to the integer nature of the feedback counter. It possesses a quantization error of +½ around Nor 
             Δ   =     1   N           
Assuming a uniform distribution of this error leads to the noise spectral density function described by
 
     
       
         
           
             
               
                 e 
                 rms 
                 2 
               
               ⁡ 
               
                 ( 
                 f 
                 ) 
               
             
             = 
             
               1 
               
                 12 
                 ⁢ 
                 
                   N 
                   2 
                 
                 ⁢ 
                 
                   f 
                   REF 
                 
               
             
           
         
       
     
     The ΔΣ modulator found in this polar transmitter shapes the quantization noise according to the transfer function
 
ΔΣ( z )=(1− z   −1 ) L  
 
where L is the order of the modulator. It in turn feeds the feedback counter, which acts a digital accumulator and reduces its noise-shaping effects. That is, the feedback counter operates in such a way that the current output phase depends on its previous output phase. As a result, the transfer function of the feedback counter or prescalar becomes
 
               P   ⁡     (   z   )       =     2   ⁢           ⁢   π   ⁢           ⁢       z     -   1         1   -     z     -   1                   
Combining the above equations shows that the noise at the output of the feedback counter equals
 
 n   2 ( f )= e   rms   2 ( f )[ΔΣ( f )] 2 [ P ( f )] 2 which simplifies to  
 
                 n   2     ⁡     (   f   )       =       1   3     ⁢           π   2         N   2     ⁢     f   REF         ⁡     [     2   ⁢     sin   ⁡     (       π   ⁢           ⁢   f       f   REF       )         ]         2   ⁢     (     L   -   1     )                 
Ultimately, this noise must be attenuated by the loop filter and PLL transfer function to avoid excessive ΔΣ noise at the output of the PLL as shown in  FIG. 3(   b ).
 
     To support wideband direct phase/frequency modulation, the fractional-N phase-locked loop adds a direct path to the VCO as shown in  FIG. 4(   a ). This modifies the VCO output to
 
ν out ( t )= A  cos [ω t+K   νCO ∫ν ctrl ( t ) dt+K   FM ∫ν FM ( t ) dt] 
 
where K VCO  and K FM  represent the sensitivity of the control port and the direct frequency modulation port, respectively. The FM signal also feeds the ΔΣ modulator and the feedback counter. This results in two paths for the FM signal as illustrated in  FIG. 4(   b ) and described by the transfer functions
 
               Δ   ⁢           ⁢   f     =             K   PD     ⁢     Z   ⁡     (   s   )       ⁢     K   V         sN   +       K   PD     ⁢     Z   ⁡     (   s   )       ⁢     K   V           ⁢   FM   ⁢           ⁢   Δ   ⁢           ⁢   f     =         sNK   FM       sN   +       K   PD     ⁢     Z   ⁡     (   s   )       ⁢     K   V           ⁢   α   ⁢           ⁢   FM             
where K PD  is the charge pump&#39;s gain, Z(s) is the impedance presented by the loop filter, K V  is the VCO&#39;s sensitivity at the tuning port, N is the value of the feedback counter, K FM  is the VCO&#39;s gain at the modulation port, and α is a scaling parameter. Ideally, these two functions combine to realize a flat response. That is, the ΔΣM path&#39;s response transitions smoothly to the VCO path&#39;s response and holds their combination at unity (0 dB). By its nature, the frequency modulation developed through the ΔΣ modulator is exact while, in contrast, the modulation formed at the VCO is sensitive to its gain K FM  and the accuracy of scaling parameter α.
 
     A key component of a direct phase/frequency modulator is a VCO such as the one shown in  FIG. 5(   a ). It uses complementary MOS devices to replenish the losses in the Le resonator. The Le resonator consists of a differential inductor, coarse-tuning capacitors (not shown), and two variable capacitance structures based on accumulation mode MOSFET devices. The accumulation-mode devices normally display an abrupt response, but impressing the large VCO signal across two back-to-back devices tends to linearize the response as shown in  FIG. 5(   b ). This is particularly important for wideband frequency modulation. 
     By design, signals applied to the control and modulation ports change the phase/frequency of the VCO output. Unfortunately, the VCO cannot discriminate between intended signals and noise. It therefore becomes important to minimize the noise as well as the sensitivity of these ports. Adding coarse-tuning capacitors to subdivide the VCO range lowers the sensitivity of the control port. Unfortunately, the nonlinear operations that formed the FM signal produce impulses as strong as one-half the FM data rate as shown in  FIG. 6 . 
     In accordance with aspects of the present invention, it is possible to realize these strong FM impulses by applying a split FM concept. This approach splits the FM signal into a coarse term FM dig  and a fine residue ΔFM where
 
 FM=FM   dig   +ΔFM  
 
In practice, the coarse term may be realized using switched capacitors similar to the coarse-tuning capacitors already found in the VCO. The fine component may then be translated to an analog signal to drive the modulation port.
 
       FIG. 7(   a ) illustrates one embodiment of a network for splitting the FM signal in accordance with aspects of the invention. As shown in  FIG. 7(   a ), the FM signal may be compared to the threshold ±TH 1  to produce a digital data stream d representing the coarse signal, an adjustment signal, and the residue analog signal ΔFM. The digital data stream may then be used to control the switched capacitors ΔC, shown in  FIG. 7(   b ), to shift the frequency of the VCO, 
               Δ   ⁢           ⁢   f     =       ±     2   3       ⁢     max   ⁡     (   FM   )               
also known as FM dig . In one embodiment, this approach may be used to advantageously reduce the residual analog signal AFM to one-third its original amplitude as shown in  FIG. 7(   c ); however, other division ratios may also be used in various embodiments. It follows that the threshold equals
 
               TH   1     =       ±     1   3       ⁢     max   ⁡     (   FM   )               
for the embodiment shown.
 
     In practice, the VCO oscillates at the resonance frequency for the LC tank described by 
                 f   c     -     Δ   ⁢           ⁢   f       =     1     2   ⁢           ⁢   π   ⁢       L   ⁡     (     C   +     Δ   ⁢           ⁢   C       )                   
where f c  represents the (un-modulated) carrier frequency and Δf corresponds to the frequency deviation or step caused by ΔC. This can be approximated by
 
               Δ   ⁢           ⁢   f     ≈       f   c     ⁡     (       Δ   ⁢           ⁢   C     C     )             
when ΔC is much smaller than C. As seen, the frequency step Δf varies with the frequency f c  and the capacitances ΔC and C. To properly split the FM signal, Δf must be adjusted or at least known.
 
     Additional threshold steps may also be used. For example, a second set of thresholds at ±TH 2  may be used to further reduce the range of ΔFM to about 1/9th its original amplitude, or to other proportionately smaller ratios. 
     Embodiments of this FM splitting approach may be used to scale the analog FM signal, easing the design of the VCO and its modulation port while preserving the FM impulses associated with wideband modulation. 
     The foregoing description, for purposes of explanation, used specific nomenclature to provide a thorough understanding of the invention. However, it will be apparent to one skilled in the art that specific details are not required in order to practice the invention. Thus, the foregoing descriptions of specific embodiments of the invention are presented for purposes of illustration and description. They are not intended to be exhaustive or to limit the invention to the precise forms disclosed; obviously, many modifications and variations are possible in view of the above teachings. The embodiments were chosen and described in order to best explain the principles of the invention and its practical applications, they thereby enable others skilled in the art to best utilize the invention and various embodiments with various modifications as are suited to the particular use contemplated. It is intended that the following claims and their equivalents define the scope of the invention.