Abstract:
A wireless communication device (e.g., a cellular telephone) includes one transceiver for voice or data communication and a global positioning system (GPS) receiver a signal for receiving a GPS signal from the GPS satellites. The GPS receiver does not receive the GPS signal when the transceiver is transmitting, so that the GPS signal receives may consist of multiple signal segments of various duration and various delays. A method is provided which combine correlations of these multiple signal segments cumulatively until a sufficiently signal-to-noise ratio is achieved to allow detection of the transmitted signal of one or more of the GPS satellites.

Description:
CLAIM OF PRIORITY 
     This application is a continuation of U.S. utility application entitled “SIGNAL DETECTOR EMPLOYING CORRELATION ANALYSIS OF NON-UNIFORM AND DISJOINT SAMPLE SEGMENTS,” having Ser. No. 09/97 1,293, filed Oct. 4, 2001 now U.S. Pat. No. 6,636,178, which is a continuation of U.S. utility application entitled “SIGNAL DETECTOR EMPLOYING CORRELATION ANALYSIS OF NON-UNIFORM AND DISJOINT SAMPLE SEGMENTS,” having Ser. No. 09/281,741, filed Mar. 30, 1999, now U.S. Pat. No. 6,304,216, both of which are entirely incorporated herein by reference. 
    
    
     BACKGROUND 
     1. Field of the Invention 
     This invention relates to the field of signal detection using correlation analysis, and more specifically, to correlation analysis in which the results of analyzing segments of samples separated in time and possibly having non-uniform lengths are combined to achieve a greater effective signal to noise ratio (SNR). 
     2. Background of the Invention 
     The Global Positioning System (GPS) is a collection of 24 earth-orbiting satellites. Each of the GPS satellites travels in a precise orbit about 11,000 miles above the earth&#39;s surface. A GPS receiver locks onto at least 3 of the satellites, and responsive, thereto, is able to determine its precise location. Each satellite transmits a signal modulated with a unique pseudo-noise (PN) code. Each PN code comprises a sequence of 1023 chips which are repeated every millisecond consistent with a chip rate of 1.023 MHz. Each satellite transmits at the same frequency. For civil applications, the frequency is known as L 1  and is 1575.42 MHz. The GPS receiver receives a signal which is a mixture of the transmissions of the satellites that are visible to the receiver. The receiver detects the transmission of a particular satellite by correlating the received signal with shifted versions of the PN code for that satellite. If the level of correlation is sufficiently high so that there is a peak in the level of correlation achieved for a particular shift and PN code, the receiver detects the transmission of the satellite corresponding to the particular PN code. The receiver then used the shifted PN code to achieve synchronization with subsequent transmissions from the satellite. 
     The receiver determines its distance from the satellite by determining the code phase of the transmission from the satellite. The code phase (CP) is the delay, in terms of chips or fractions of chips, that a satellite transmission experiences as it travels the approximately 11,000 mile distance from the satellite to the receiver. The receiver determines the code phase for a particular satellite by correlating shifted versions of the satellite&#39;s PN code with the received signal after correction for Doppler shift. The code phase for the satellite is determined to be the shift which maximizes the degree of correlation with the received signal. 
     The receiver converts the code phase for a satellite to a time delay. It determines the distance to the satellite by multiplying the time delay by the velocity of the transmission from the satellite. The receiver also knows the precise orbits of each of the satellites. The receiver uses this information to define a sphere around the satellite at which the receiver must be located, with the radius of the sphere equal to the distance the receiver has determined from the code phase. The receiver performs this process for at least three satellites. The receiver derives its precise location from the points of intersection between the at least three spheres it has defined. 
     The Doppler shift (DS) is a frequency shift in the satellite transmission caused by relative movement between the satellite and the receiver along the line-of-sight (LOS). It can be shown that the frequency shift is equal to 
           v   LOS     λ     ,       
 
where ν LOS  is the velocity of the relative movement between the satellite and receiver along the LOS, and λ is the wavelength of the transmission. The Doppler shift is positive if the receiver and satellite are moving towards one another along the LOS, and is negative if the receiver and satellite are moving away from one another along the LOS.
 
     The Doppler shift alters the perceived code phase of a satellite transmission from its actual value. Hence, the GPS receiver must correct the satellite transmissions for Doppler shift before it attempts to determine the code phase for the satellite through correlation analysis. 
     The situation is illustrated in  FIG. 1 , which shows a GPS receiver  200  and three GPS satellites  201   a ,  201   b , and  201   c . Each satellite  201   a ,  201   b ,  201   c  is transmitting to the GPS receiver  200 . Satellite  201   a  is moving towards the GPS receiver  200  along the LOS at a velocity ν a   + ; satellite  201   b  is moving away from the GPS receiver  200  along the LOS at a velocity ν b   − ; and satellite  201   c  is moving away from the GPS receiver  200  along the LOS at a velocity ν c   − . Consequently, assuming a carrier wavelength of λ, the transmission from satellite  201   a  will experience a positive Doppler shift of 
           v   a   +     λ     ;       
 
the transmission from satellite  201   b  will experience a negative Doppler shift of 
           v   b   -     λ     ;       
 
and the transmission from satellite  201   c  will experience a negative Doppler shift of 
           v   c   -     λ     .       
 
     The GPS receiver functions by sampling a finite portion of the received signal and then processing the samples. Typically, external constraints limit the size and occurrence of the sampling period. For example, in the case of a mobile wireless phone integrated with a GPS receiver, the sampling window should be limited to those periods in which the phone is not transmitting. The purpose is to avoid interference between the transmitter and the GPS receiver. 
     The problem is that the signal to noise ratio of the received signal over a finite sampling window may not be sufficient to detect the presence and range of the satellites. For example, the signal may be such that there is no correlation value for a particular set of hypotheses which is significantly larger than the correlation values resulting from the other hypotheses tested. 
     Moreover, it is difficult to combine segments of samples captured over different periods of time because each is subject to a different code phase which must be accounted for before the segments can be combined, and these code phases are unknown. In an effort to increase the signal to noise ratio of the received signal, prior art receivers are thus required to either forgo operation during times in which the received signal is weak, or to extend the sampling period beyond the limits imposed by external constraints. In certain applications, such as the case of a GPS receiver integrated with a mobile wireless phone, extension of the sampling window is not usually feasible since it would subject the received signal to unacceptable interference from the phone&#39;s transmitter. In such applications, the practical effect is to forego operation of the GPS receiver when the received signal is weak. Such occurrences are frequent because of the approximately 11,000 distance traveled by the GPS satellite transmissions, and because of the noise to a particular satellite represented by the other satellite&#39;s transmissions. 
     Consequently, there is a need for a signal detector which overcomes the disadvantages of the prior art. Similarly, there is a need for a GPS receiver which overcomes the disadvantages of the prior art. 
     RELATED APPLICATIONS 
     This application is related to U.S. patent application Ser. No. 09/145,055, filed Sep. 1, 1998, and entitled “DOPPLER CORRECTED SPREAD SPECTRUM MATCHED FILTER,” now U.S. Pat. No. 6,044,105, and U.S. patent application Ser. No. 09/281,566, filed on Mar. 30, 1999 and entitled “SIGNAL DETECTOR EMPLOYING COHERENT INTEGRATION,” now U.S Pat. No. 6,577,271 both of which are owned in common by the assignee hereof, and both of which are hereby fully incorporated by reference herein as though set forth in full. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  illustrates an example environment for operation of a GPS receiver. 
         FIG. 2  illustrates an embodiment of a method of operation of a GPS receiver in accordance with the subject invention. 
         FIGS. 3A–3C  illustrate an embodiment of a method of operation of a GPS receiver in accordance with the subject invention. 
         FIG. 4  illustrates an embodiment of a GPS receiver in accordance with the subject invention. 
         FIG. 5  illustrates an embodiment of a matched filter in accordance with the subject invention. 
         FIG. 6  illustrates the data structures output by one embodiment of a matched filter in accordance with the subject invention. 
         FIG. 7  illustrates data flows in one embodiment of a GPS receiver in accordance with the subject invention. 
         FIGS. 8A–8B  illustrates an embodiment of a method of operation of a GPS receiver in accordance with the subject invention. 
         FIG. 9  illustrates an embodiment of a method of operation of a GPS receiver in accordance with the subject invention. 
         FIG. 10  is a timing diagram for combining multiple correlation arrays in accordance with one implementation of the subject invention. 
         FIGS. 11A–11C  illustrates an embodiment of a method of operation of a matched filter in accordance with the subject invention. 
         FIG. 12  is a block diagram of an embodiment of a signal detector in accordance with an embodiment of the subject invention. 
         FIG. 13  is a flowchart illustrating an embodiment of a method of operation of the signal detector of  FIG. 12 . 
     
    
    
     SUMMARY OF THE INVENTION 
     In accordance with the purpose of the invention as broadly described herein, there is provided a signal detector configured to combine the results of performing correlation analysis on segments of samples of the received signal that may have non-uniform lengths and that may have been obtained over different and non-overlapping periods of time. In one embodiment, the segments are obtained during sampling windows of arbitrary length and at arbitrary times, and the results of processing the segments are successively combined until a threshold signal to noise ratio has been achieved. 
     In one embodiment, the signal detector is part of a GPS receiver. In this embodiment, the GPS receiver comprises a radio frequency (RF) receiver, sampling circuitry, timing circuitry, offset measurement circuitry, a PN code generator, a matched filter, and a GPS processor. The RF receiver demodulates the received signal to obtain a baseband signal. The sampling circuitry provides, responsive to timing signals produced by the timing circuitry, a segment of samples of the baseband signal taken over a defined sampling window. The matched filter processes the segment of samples in accordance with a plurality of PN code, Doppler shift, and code phase hypotheses. 
     In one implementation, for each segment of samples, the matched filter outputs correlation data derived by correlating various combinations of PN code, Doppler shift and code phase hypotheses with the segment of samples. According to this implementation, the correlation data can be grouped into groupings which correspond to various combinations of specific hypotheses and ranges of hypotheses. In on implementation example, the correlation data comprises a plurality of arrays, wherein each array corresponds to a PN code hypothesis, each row of an array corresponds to a Doppler shift hypothesis, each column of an array corresponds to a code phase hypothesis, and each entry in the array is a measure of the degree to which the combined PN code, Doppler shift, and code phase hypothesis corresponding to the entry correlates to the samples. 
     The PN code generator generates the PN code hypotheses which are provided as inputs to the matched filter. In one embodiment, the Doppler shift hypotheses are generated internally within the matched filter. The GPS processor sends out data capture commands to the sampling circuitry and the matched filter directs the sampling circuitry to capture a segment of samples, and directs the matched filter to process the segment of samples. The GPS processor also, responsive to timing signals generated locally by the GPS radio receiver, generates frame marks which are input to the offset measurement circuitry. The offset measurement circuitry determines the offset between the time that a data capture command is issued by the GPS processor and the timing of the next frame mark. This information is provided to the GPS processor for use in combining the results of processing different segments of samples. 
     In one embodiment, cumulative correlation data is maintained. When the correlation data for a new segment of samples is obtained, the new correlation data is combined with the cumulative correlation data. According to this embodiment, the GPS processor accumulates the correlation data for a particular satellite until a threshold signal to noise ratio is achieved. In one implementation, the data for a particular satellite is maintained until the presence and range of the satellite can be determined. 
     The correlation data for different segments of samples are combined using an algorithm that allows for the differing code phases between the segments to be accounted for even through the actual code phases are unknown. In one embodiment, in which the correlation data for a given segment of samples comprises a plurality of arrays, wherein each array in the plurality corresponds to a particular PN code hypothesis, and each row of an array corresponds to a particular Doppler shift hypothesis, cumulative correlation arrays are initialized with the correlation arrays for the first segment. Then, the correlation arrays for the second segment are combined with the cumulative correlation arrays one array at a time, and within an array, one row at a time. The process continues indefinitely and ceases for a particular satellite (PN code) when the presence and range of the satellite is accurately detected. 
     The code phase difference corresponding to a row is derived from the total time offset between the start of the first segment and the start of the second segment, and the Doppler shift hypothesis corresponding to the row. 
     In one implementation example, the code phase difference is determined from the following equation:
 
Δ CP =[( F   PN   +D )×Δ T ]modulo 1 mS
 
in which ΔCP is the code phase difference, F PN  is the nominal PN rate of 1.023 MHz, D is the Doppler shift hypothesis corresponding to the array, and ΔT is the time offset between the beginning of the sampling period for the first segment, and that for the second segment. The value of 1 mS is the period of the PN codes. Since the codes are periodic, the code phase difference of interest is the fractional part of the code period. This is reflected in the foregoing equation through the modulo 1 mS operation.
 
     In this implementation example, the first and second segments each span at least a frame, and begin at a time which is offset from a frame mark. The offsets for the first and second segment, OS 1  and OS 2 , respectively, are each determined from the offset measurement circuitry. According to the implementation example, the offsets are used to determine ΔT in the foregoing equation as follows:
 
Δ T =( T   n   −T   2   +OS   1   −OS   2 )× S 
 
     In this latter equation, T n  is the timing of the frame mark which occurs at the beginning of the second segment, and from which OS 2  is defined. Similarly, T 2  is the timing of the frame mark which occurs at the beginning of the first segment, and from which OS 1  is defined. The variable S is the error in the local time base generated by a local crystal oscillator in the GPS receiver in relation to the time base defined at the GPS satellites. The GPS receiver determines this error and corrects for it in the foregoing equation. 
     This code phase difference is then used to combine the rows for the correlation arrays. In the implementation example, the row of correlation data for the second segment is circularly shifted by the code phase difference. Then, the shifted row is added to the corresponding row of cumulative correlation data, one data element at a time. This process is then repeated for each of the rows of each of the arrays for the second segment. The result is a plurality of cumulative correlation arrays which combine the results of the first and second segments. 
     The correlation arrays for additional segments are combined with the cumulative correlation arrays in the foregoing manner. For a given satellite, the process of combining segments continues until the signal to noise ratio is sufficient to permit the presence and range of the satellite to be accurately determined. 
     Related methods of operation and computer readable media are also provided. 
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     A block diagram of one embodiment of a signal detector in accordance with the subject invention is illustrated in  FIG. 12 . As illustrated, the signal detector comprises a receiver  2000  configured to receive segments of a signal. The segments may be of non-uniform-length and may be disjoint and separated by arbitrary periods of time. The signal may comprise a signal of interest perturbed by noise. Alternatively, in accordance with a spread spectrum environment, the signal may comprise the combination of multiple signals of interest each encoded using one of a plurality of pseudo-noise (PN) codes. In this case, other signals appear as noise to a particular signal of interest. 
     A hypothesis generator  2001  generates a plurality of hypotheses about the signal of interest. A correlator  2003  receives the plurality of hypotheses from hypothesis generator  2001 , and the segment received by receiver  2000 , and responsive thereto, generates correlation data representative of the correlation between the received segment and the plurality of generated hypotheses. 
     The correlation data is provided to combiner  2003  which combines the correlation data with cumulative correlation data accumulated for previous segments received by receiver  2000 . In one embodiment, the combiner  2003  first performs an adjustment on the correlation data so that it is combinable with the cumulative correlation data. Once the data has been combined, combiner  2003  determines whether the cumulative correlation data is sufficient to permit a parameter of the signal of interest to be detected. If so, an output is provided on signal line  2004  signaling detection of the parameter of the signal of interest. If not, a signal is provided to the receiver on signal line  2006  indicating the need for additional segments. Optionally, the foregoing process iterates until the parameter of the signal of interest is detected (or a time-out condition is detected). 
     A flowchart of one embodiment of a method of operation of a signal detector in accordance with the subject invention is illustrated in  FIG. 13 . As illustrated, the process begins at step  2011 , in which a segment of the type described earlier is received. Then, in step  2007 , a plurality of hypotheses are generated for testing. In step  2008 , correlation data measuring the level of correlation between the received sample and the generated hypotheses is derived. 
     In step  2009 , the correlation data from step  2009  is combined with cumulative correlation data which may be present from correlation analysis performed on previous segments. In one embodiment, the correlation data from step  2009  is first adjusted or altered so that it is combinable with the cumulative correlation data. If no such data is present, the cumulative correlation data is initialized with the correlation data derived in step  2008 . 
     In step  2010 , a determination is made whether the cumulative correlation data permits accurate and reliable detection of a desired parameter of the signal of interest. If so, the process ends. If not, a jump is made to step  2011  and the process repeats itself from this point with a new segment. Optionally, the process iterates until the desired parameter of the signal of interest is detected (or a time-out condition occurs). 
     It is contemplated that the foregoing signal detector can be beneficially employed in a variety of applications, such as in a GPS receiver. One embodiment of a GPS receiver in accordance with the subject invention is illustrated in  FIG. 4 . As shown, the receiver comprises a radio frequency (RF) receiver  300 , sampling circuitry  308 , timing circuitry  307 , offset measurement circuitry  305 , a PN code generator  312 , a matched filter  310 , and a GPS processor  303 . The RF receiver section  300  demodulates the received signal to obtain a baseband signal which is provided to the sampling circuitry  308  over signal line  302 . The sampling circuitry  308  provides, responsive to timing signals produced by the timing circuitry  307 , a segment of samples of the baseband signal taken over a defined sampling window. The segment of samples is provided to the matched filter  310  over signal line  309 . The matched filter  310  processes the segment of samples in accordance with a plurality of PN code, Doppler shift, and code phase hypotheses. 
     In one implementation, as illustrated in  FIG. 6 , for each segment of samples, the matched filter outputs a plurality of correlation arrays  600 ,  601 ,  602  of data derived from the segment of samples. According to this implementation, each array  600 ,  601 ,  602  corresponds to a PN code hypothesis, PN 1 , PN 2 , . . . PNr, each row of an array  600 ,  601 ,  602  corresponds to a Doppler shift hypothesis, DS  1 , DS 2 , . . . DSm, each column of an array  600 ,  601 ,  602  corresponds to a code phase hypothesis, CP 1 , CP 2 , . . . CPn, and each entry in an array  600 ,  601 ,  602  is a measure of the degree to which the combined PN code, Doppler shift, and code phase hypothesis corresponding to the entry correlates to the samples. Thus, in  FIG. 5 , numeral  500  identifies a correlation array corresponding to PN code hypothesis PN 1 ; numeral  501  identifies a correlation array corresponding to PN code hypothesis PN 2 ; and numeral  502  identifies a correlation array corresponding to PN code hypothesis PNr. 
     The PN code generator  312  generates the PN code hypotheses which are provided as inputs to the matched filter  310  over signal line  315 . In one embodiment, the Doppler shift hypotheses are generated internally within the matched filter. The GPS processor  303  issues data capture commands on signal line  314  to the sampling circuitry  308  and the matched filter  310  directs the sampling circuitry  308  to capture a segment of samples, and directs the matched filter  310  to process the segment of samples. The GPS processor  303  also, responsive to timing signals generated locally by the GPS radio receiver  300  and received over signal line  301 , generates frame marks which are input to the offset measurement circuitry  305  over signal line  304 . In one implementation, the timing signals generated by the RF receiver  300  are generated by a local oscillator within the RF receiver. In one implementation, the timing signals define a local time base which is related to the time base maintained by the GPS satellites. 
     The offset measurement circuitry  305  determines the offset between the time that a data capture command is issued by the GPS processor and the time of the next frame mark. This information is provided to the GPS processor  303  for use in combining, in a manner to be described subsequently, the results of processing different segments of samples. 
     In one embodiment, the correlation arrays for a segment are grouped by PN code hypothesis, and by Doppler shift hypothesis for a given PN code hypothesis. The result is that each grouping corresponds to a particular combination of PN code hypothesis and Doppler shift hypothesis. In this embodiment, the correlation arrays are combined one grouping at a time. According to this implementation, the GPS processor  303  receives these groupings over signal line  313 , and cumulatively combines these correlation arrays as additional segments are captured. For a particular satellite, the combination process continues until a threshold signal to noise ratio is obtained for the satellite. In one implementation example, the correlation arrays corresponding to a satellite are combined until the presence and range of the satellite can be accurately determined. Typically, this occurs when the correlation data for a particular set of hypotheses is significantly greater than the correlation data for alternative hypotheses. 
     The correlation arrays derived from different segments of samples are combined using an algorithm that accounts for the differing code phases between the segments even though the actual code phases are unknown. In one embodiment, a plurality of cumulative correlation arrays are maintained, which are initially set equal to the correlation arrays derived from a first segment of samples. The correlation arrays for a second segment of samples are then combined with the cumulative correlation arrays one row at a time. According to this embodiment, the code phase difference corresponding to a row is derived from the total time offset between the start of the first segment and the start of the second segment, and the Doppler shift hypothesis corresponding to the row. A circular shift of the row for the second segment is then performed, with the amount of the shift being derived from the code phase difference which has been determined. The entries in the shifted row are then added to the corresponding entries of the corresponding row in the cumulative arrays. 
     The foregoing combining process is then repeated for additional data in the correlation arrays for the second segment. In one implementation, the process is repeated for each of the rows inn each of the arrays remaining in the correlation data for the second segment. Moreover, once the data for the second segment has been exhausted, the data for additional segments is combined with the cumulative arrays in the foregoing manner until, for a given satellite, the signal to noise ratio is sufficient to permit the presence and range of the satellite to be accurately determined. 
     The situation is illustrated in  FIG. 7 . Numeral  600  identifies one of the arrays in the plurality of cumulative correlation arrays which are maintained. The particular array which is identified is the one corresponding to PN code hypothesis PNi. These arrays are assumed to have been initially set equal to the correlation arrays derived from a first segment of samples. Numeral  601  identifies the corresponding one of the arrays in the plurality of correlation arrays which are derived from a second set of samples. Again, this array corresponds to the PN code hypothesis PNi. In one implementation, the first segment of samples is obtained during a first time period, and the second set of samples is obtained during a subsequent non-consecutive time period. 
     The row  603  from array  601  is combined with the row  602  from array  600  in the manner shown. Both rows are assumed to correspond to the same Doppler shift hypothesis, DPi. Row  602  is obtained as indicated by identifying numeral  604 . In addition, row  603  is obtained as indicated by identifying numeral  605 . As indicated by identifying numeral  607 , the code phase difference, ΔCP, defining the code phase difference between the start of the first and second segments is determined. The row  603  obtained from array  601  is then circularly shifted by an amount derived from ΔCP. Such is indicated by identifying numeral  606  in  FIG. 7 . The entries in the shifted row are then added to corresponding entries in row  602 . Such is indicated by identifying numeral  608 . The result is then stored in array  600  in lieu of row  602 . Such is indicated by identifying numeral  609 . 
     In one implementation example, the code phase difference is determined from the following equation:
 
Δ CP =[( F   PN   +D )×Δ T ]modulo 1 mS  (1)
 
,in which ΔCP is the code phase difference, F PN  is the nominal PN rate of 1.023 MHz, D is the Doppler shift hypothesis corresponding to the row, and ΔT is the time offset between the beginning of the sampling period for the first segment, and that for the second segment. The value of 1 mS is the period of the PN codes. Since the codes are periodic, the code phase difference of interest is the fractional part of the code period. This is reflected in the foregoing equation through the modulo 1 mS operation.
 
     In this implementation example, the first and second segments each span at least a frame, and begin at a time which is offset from a frame mark. The offsets for the first and second segments, OS 1  and OS 2 , respectively, are each determined from the offset measurement circuitry  306 . According to the implementation example, the offsets are used to determine ΔT in the foregoing equation as follows:
 
Δ T =( T   n   −T   2   +OS   1   −OS   2 )× S   (2)
 
     In this latter equation, T n  is the timing of the frame mark which occurs at the beginning of the second segment, and from which OS 2  is defined. Similarly, T 2  is the timing of the frame mark which occurs at the beginning of the first segment, and from which OS 1  is defined. The variable S is the error in the local time base generated by a local crystal oscillator in the RF receiver  300  in relation to the time base defined at the GPS satellites. The GPS receiver determines this error and corrects for it in the foregoing equation. 
     The situation is illustrated in  FIG. 10 . Local time base  900  is divided into frames by frame marks T 1 , T 2 , . . . T n . In one implementation example, the duration of a frame is 20 mS. Upon detecting the occurrence of frame mark T 1 , identified in the figure with numeral  901 , GPS processor  303  issues a command, identified with numeral  904 , to capture a first segment of samples. The capture of the first segment of samples begins at time  910 , and the segment itself is identified in the figure with numeral  908 . As indicated, the first set of samples is assumed to span at least one frame. The offset measurement circuitry  305  detects the next occurrence of a frame mark, T 2 , identified in the figure with numeral  902 , and, responsive thereto, determines the offset, OS 1 , between the time  910  that capture of the segment began, and the time  902  of the frame mark T 2 . 
     The segment of samples  908  is typically representative of the combination of transmissions from multiple ones of GPS satellites, each of which has a different code phase as measured at the receiver. In relation to a particular satellite of interest, the segment  908  will have a specific code phase, CP 1 , which is identified in the figure with numeral  906 . 
     After the occurrence of a number of additional frame marks, T 3 , . . . T n−1 , the GPS processor  303  issues another command to capture a segment of samples. This second command is identified in the figure with numeral  905 , and is assumed to occur after the GPS processor  303  has detected the occurrence of frame mark T n−1 . The capture of the second segment, which is identified with numeral  909 , begins at time  911 . Again, the second segment is assumed to span at least one frame. The next frame mark, T n , occurs at time  903 . The offset measurement circuitry  305  measures the offset, OS 2 , between the beginning  911  of the second segment, and the occurrence  903  of the next frame mark T n . This second frame is assumed to have a particular code phase, CP 2 , which is defined in relation to a particular satellite. The time ΔT in the foregoing equations is the difference in time between the beginning  911  of the second segment, and the beginning  910  of the first segment, and the value ΔCP in the foregoing equations is the difference in code phase between the code phase for the first segment, CP 1 , and that for the second segment, CP 2 . Significantly, this value is determined from the foregoing equations even though the underlying code phases CP 1  and CP 2  are unknown. 
       FIG. 5  illustrates one embodiment of matched filter  310  in  FIG. 4 . Compared to  FIG. 4 , like elements in  FIG. 5  are referenced with like identifying numerals. As illustrated, this embodiment of the matched filter comprises random access memory (RAM)  400  which is configured to receive samples from sampling circuitry  308  over signal line  309 . In one implementation example, a frame is 20 mS in duration, and the RAM  400  is configured to receive one 20 mS frame of samples at a time. According to this implementation example, each 20 mS frame of samples comprises 40920 samples, obtained by sampling the baseband signal at a nominal sampling rate of 20.46 MHz, and then performing decimation filtering. 
     Also provided is Doppler correction circuitry  403 , and Doppler shift generator  401 . The RAM  400  is configured to provide to Doppler correction circuitry  403  over signal line  404  at least a portion of a frame of samples stored therein at a time. In one implementation, the frame of samples stored in RAM  400  is divided into subframes, and is provided to Doppler correction circuitry  403  one subframe at a time. In one implementation example, the duration of a subframe is equal to the period of the PN codes, which is currently set to 1 mS. In this implementation example, each subframe of samples comprises 2046 samples, with each sample represented as a complex number having an in-phase (I) and a quadrature (Q) component which are typically expressed in combination as I+jQ. According to this implementation example, each component, whether I or Q, is represented by 2 bits, and can take on any one of the discrete values −1, 0, or +1. 
     Doppler shift generator  401  generates a plurality of Doppler shift hypotheses which are provided to Doppler correction circuitry  403  over signal line  405  one hypothesis at a time. In one implementation example, the Doppler shift generator  401  generates Doppler shift hypotheses in the range of ±62,000 Hz, to allow for additional inaccuracy in the local time base that is not corrected by the input sampling process. 
     Sample register  406  is also provided. Doppler correction circuitry  403  receives a subframe of samples from RAM  400  over signal line, and a Doppler shift hypothesis from Doppler shift generator  401 , and, responsive thereto, generates a corrected subframe of samples which are stored in sample register  406 . Additional detail about this procedure is available in U.S. patent application Ser. No. 09/145,055, filed Sep. 1, 1998, entitled “DOPPLER CORRECTED SPREAD SPECTRUM MATCHED FILTER,” now U.S. Pat. No. 6,044,105 previously incorporated by reference herein as though set forth in full. In one implementation example, each corrected subframe of samples continues to comprise 2046 complex samples, each having I and Q components. Each of the samples in this implementation example can take on any one of the discrete values −2, −1, 0, +1, and +2, which are represented by 3 bits for each of I and Q. 
     PN code register  415  is provided to store the current PN code hypothesis provided by PN code generator  312  over signal line  315 . In one implementation, each PN code hypothesis represents one period of a PN code. In one implementation example, the PN code period is 1 mS, and each PN code hypothesis represents 1023 chips which repeats every 1 mS, representing a chip rate of 1.023 MHz. In this implementation example, the PN code register is configured to store 1023 chips at a time. 
     As indicated by signal line  414 , the PN code register is capable of being circularly shifted by an amount which corresponds to a code phase delay hypothesis. In the implementation example which has been discussed, in which the period of a PN code is 1023 chips, the value of the code phase delay can range from 0 to 2045 half chip increments. The PN code register is configured in this implementation example to be circularly shifted by any number or fraction of chips which correspond to a code phase delay hypothesis under consideration. 
     Sum of products circuitry  407  is also provided. This circuitry is configured to form the integration of the product between the subframe of corrected samples stored in sample register  406  and the PN code hypothesis stored in the PN code register  414 . 
     In the implementation example discussed earlier in which the subframe of samples stored in sample register  406  comprises 2046 samples, each having I and Q components, and the PN code hypothesis stored in PN code register  415  comprises 1023 chips, a correspondence is present between two of the samples in the sample register  406 , and one of the chips in PN code register  415 . The I and the Q components of each of the two samples is multiplied by the corresponding PN chip. Then, the sum of the I component products is determined, and the sum of the Q component products is separately determined. The sum of the I component products is output on signal line  408 , and the sum of the Q component products is output on signal line  409 . 
     In equation form, the function of the sum of products circuitry  407  in this implementation example can be represented as follows: 
             SI   =       ∑     i   =   1     1023     ⁢       CHIP   i     ×     (       I   i   1     +     I   i   2       )                 (   3   )               SQ   =       ∑     i   =   1     1023     ⁢       CHIP   i     ×     (       Q   i   1     +     Q   i   2       )                 (   4   )             
 
where CHIP i  is the ith chip in the PN code hypothesis, I i   l  is the I component of the first of the two samples corresponding to CHIP i , I i   2  is the I component of the second of the two samples corresponding to CHIP i , Q i   1  is the Q component of the first of the two samples corresponding to CHIP i , and Q i   2  is the Q component of the second of the two samples corresponding to CHIP i .
 
     Sqrt. of sum of squares circuitry  410 , adder  411 , and RAM  413  are also provided. The sqrt. of sum of squares circuit  410  is configured to receive the sum of the I component products (SI) and the sum of the Q component products (SQ) on signal lines  408  and  409  respectively, and to determine the square root of the sum of the squares of these two values. In equation form, the circuit computes the value:
 
 SS =√{square root over (( SI ) 2 +( SQ ) 2 )}{square root over (( SI ) 2 +( SQ ) 2 )}.  (5)
 
     If the subframe under consideration and stored in sample register  406  is the first subframe for the frame of interest, the foregoing value is stored in an array entry in RAM  413  corresponding to the combination of the PN code, Doppler shift, and code phase hypotheses under consideration. The arrays are of the same format as those depicted in  FIG. 6  and will eventually become the correlation arrays for the current PN hypothesis. 
     If the subframe under consideration and stored in sample register  406  is not the first subframe analyzed for the frame of interest, there may already be a value derived from a previous subframe stored in RAM  413  in the entry corresponding to the combination of the PN code, Doppler shift, and code phase hypotheses under consideration. In this case, the SS value determined above is added by adder  411  with the previously stored value which is provided to the adder  411  over signal line  412 . The result is then stored in lieu of the previously stored value in the array entry corresponding to the combined PN code, Doppler shift, and code phase hypotheses. 
     The next code phase hypothesis is then selected, and the PN code register  407  circularly shifted in accordance with the selected code phase hypothesis. The foregoing process is then repeated. This process continues for each of the code phase hypotheses which are desired to be tested. In one implementation, 2046 code phases are tested for each 1 mS subframe, corresponding to the repetition period of the PN codes. In this implementation, the code phase hypotheses which are tested range from 0 to 2045 half-chip increments, and the next code phase hypothesis is selected simply by circularly shifting the PN code register  415  by one-half chip. 
     The matched filter of  FIG. 5  is configured to perform the foregoing tasks for each of the subframes of the frame stored in the RAM  400 . When this process has been completed, correlation arrays of the form shown in  FIG. 6  are present in RAM  313 . These correlation arrays are provided to the GPS processor  303  over signal line  313 . GPS processor  303  combines these correlation arrays with correlation arrays derived from a previous segment in the manner described previously. 
     A first embodiment of an overall method of operation of the subject invention is illustrated in  FIG. 2 . As illustrated, the process is iterative, and each cycle of the process begins at step  1 . There, a segment of samples is received. In one embodiment, the segment size is variable from cycle to cycle. In another embodiment, the segment size is fixed. In one implementation example, the segment size is 20 mS. 
     In step  2 , the PN code, Doppler shift, and code phase hypotheses to be tested are determined. In one embodiment, these hypotheses are variable from cycle to cycle. In another embodiment, these hypotheses are fixed. In one implementation, the PN code hypotheses to be tested are determined responsive to the satellites which have already been successfully detected. If a satellite has previously been successfully detected, then its PN code is removed from the set of PN code hypotheses to be tested. Similarly, the set of Doppler hypotheses to be tested may be reduced based on the Doppler values at which previous satellites have been detected. 
     In step  3 , correlation analyses are performed on the segment of samples in accordance with the set of hypotheses to be tested as determined in step  2 . The result is correlation data which measures the level of correlation between various combinations of the tested hypotheses and the segment of samples. In one implementation example, the correlation data is computed using equations (3), (4), and (5) above. 
     In step  6 , this data is selectively combined with correlation data which has been accumulated from previous segments. In one embodiment, this step involves adjusting for the code phase difference between the current segment and the previous segments. In one implementation, this step involves combining correlation arrays row by row after shifting at least one of the rows by the code phase difference which has been determined. In one implementation example, the code phase differences are computed in accordance with equations (1) and (2) above. 
     In step  7 , a determination is made whether the data which has been accumulated permits detection of the presence and range of a satellite. If so, then step  8 , in which the range of the satellite is determined, is performed. In one implementation, this step involves determining the range from the code phase hypothesis which has the highest correlation with the samples which have been accumulated. If not, a jump is made back to step  1 , whereupon the process repeats itself with a new segment of samples. 
     A second embodiment of an overall method of operation of the subject invention is illustrated in  FIGS. 3A–3C . Again, the process is iterative, and each cycle of the process begins with step  100 . According to this process, it is assumed that a time base exists which is divided up into successive periods known as epochs. 
     In step  100 , the process loops until the beginning of an epoch is detected. In one implementation, an epoch is a frame period. In one implementation example, an epoch is a 20 mS frame period. In the implementation which is the subject of the timing diagram of  FIG. 10 , epochs are the time periods marked by T 0 , T 1 , T 2 , . . . , T n , T n+1 , etc. In the scenario depicted in  FIG. 10 , numeral  901  identifies the beginning of a new epoch which is represented by T 1 . 
     Once the beginning of an epoch is detected, step  101  is performed. In step  101 , the storage of receiver samples is initiated before the commencement of the next epoch. In the scenario depicted in  FIG. 10 , the initiation of the new epoch at T 1  generates an interrupt which causes the GPS processor  303  to issue a command to begin data capture. Such a command is identified in  FIG. 10  with numeral  904 . The command is such that data capture begins before the beginning of the next epoch. With reference to  FIG. 10 , the next epoch is identified with numeral  902 , and is represented by T 2 . In accordance with the method of FIGS.  3 A– 3 B-, the capture of data samples from the receiver is commenced at the time identified with numeral  910 , which is before the commencement of the next epoch. 
     When the next epoch is detected, a counter begins counting. This is represented in  FIG. 3A  by step  102  which is performed after step  101 . In one implementation example, the counter is part of the offset measurement circuitry  305  in  FIG. 4 . 
     Step  103  is then performed. In step  103 , the system loops until the next epoch is detected. In the scenario depicted in  FIG. 10 , the next epoch is T 2 , and is identified with numeral  902 . Once the next epoch has been detected, step  104  is performed. 
     In step  104 , the counter is stopped, and the contents of the counter used to derive the time offset OS I  between the initiation of data capture and the beginning of the next epoch. In the scenario depicted in  FIG. 10 , this value is represented as CP 1 , and is the time offset between the initiation of the capture of data segment  908  at the time identified with numeral  910 , and the time identified with numeral  902 , which is the beginning of epoch T 2 . 
     Step  105  is then performed. In step  105 , the system loops until data capture has been completed. In one implementation, the completion of data capture is detected by GPS processor  303  either by an interrupt which is generated or through a polling method. In the scenario depicted in  FIG. 10 , the completion of the capture of data samples  908  is identified with numeral  912 . 
     Upon the completion of data capture, step  106  is performed. In step  106 , the contents of the counter, which defines OS I , is saved and associated with the captured data samples. In one implementation example, the GPS processor  303  reads the contents of the counter maintained in the offset measurement circuitry  305 , and defines this value as CP 1 . The GPS processor  303  then saves this value and associates it with the data samples  908 . 
     The system then generates correlation data which describes the degree of correlation between the PN code, Doppler shift, and code phase hypotheses which are desired to be tested. In step  107 , the PN code hypothesis to be tested is selected, and in step  108 , the Doppler shift hypothesis to be tested in selected. In one implementation, the selected PN code hypothesis is generated by PN code generator  312 , and then stored in PN code register  415 . In addition, the Doppler shift hypothesis is generated by Doppler shift generator  401 . 
     Step  109  is then performed. In step  109 , the receiver samples which have been captured are corrected for the selected Doppler shift hypothesis. In one implementation, the receiver samples are stored in RAM  400 , and Doppler correction circuitry  403  corrects these samples for the selected Doppler shift hypothesis one subframe at a time. Additional detail regarding this implementation is available in U.S. Ser. No. 09/145,055, previously incorporated herein by reference. 
     Step  110  is then performed. In step  110 , a cross-correlation analysis is then performed between the adjusted data from step  109 , and the PN code hypothesis which has been selected. According to this step, the integral of the cross-product between the adjusted receiver samples and the PN code, shifted in accordance with a selected code phase hypothesis, is determined for each of a plurality of code phase hypotheses. In one implementation, 2046 code phase hypotheses are tested. These code phase hypotheses are generated by circularly shifting the selected PN hypothesis by a shift value ranging from 0 to 2045 half-chip increments. In one implementation example, this step is performed one subframe at a time. In this implementation, the integration values for the various subframes for a given PN code, Doppler shift and code phase hypothesis are simply added together to obtain the integration value for the frame. In one implementation, this step is performed by the matched filter of  FIG. 5 . In one implementation example, this step is performed in accordance with equations (3), (4), and (5) above. 
     When step  110  has been completed, step  111  is performed. In step  111 , the correlation data obtained in step  110  is combined with any previous correlation data for the same PN code/Doppler shift hypotheses in a manner to be described subsequently. Of course, if this is the first set of correlation data obtained for the given PN code/Doppler shift hypotheses, these values are simply stored. In one implementation example, the completion of the cross-correlation analysis of step  110  is detected by GPS processor  303  through either an interrupt or a polling method. In one implementation example, the correlation data for a segment is combined with any previous correlation data using equations (1) and (2) above. 
     In step  112 , a determination is made whether there are any more Doppler shift hypotheses to be tested. If so, a jump is made back to step  108 , and the process is repeated beginning at this point. If not, step  113  is performed. 
     In step  113 , a determination is made whether there are any more PN code hypotheses to be tested. If so, a jump is made back to step  107 , and the process is repeated beginning at this point. If not, step  114  is performed. 
     In step  114 , a determination is made whether there are any more data segments to be captured for combination with the results which have been accumulated. If so, a jump is made back to step  100 , and the process is repeated beginning at this point. If not, the process terminates. 
     Typically, as the process iterates, there will be fewer PN code and Doppler shift hypotheses to test on each pass. This reduction occurs because some PN codes (satellites) will have been detected and because the range of Doppler uncertainty that needs to be tested may be reduced based on observing the Doppler values at which previous satellites have been detected. 
     Also, it should be appreciated that the parameters which are collected for each set of data samples change. For example, with reference to  FIG. 10 , the first pass is associated with the parameters OS 1 , T 2 , and CP 1 . The second pass, however, is associated with the parameters T n , OS 2 , and CP 2 . 
     It should also be appreciated that the data capture process is not limited to a fixed 20 mS epoch or segment length. In general, it can be adjusted on each pass to a variable length. In one implementation example, it can adjusted on each pass to a variable length of from 1 to 20 mS, in 1 mS increments. In other implementation examples, longer periods are possible simply by increasing the amount of RAM  400 . 
     Upon the completion of the process of  FIGS. 3A–3C , in one implementation, the GPS processor  303  has stored within its memory a plurality of correlation arrays, with each array corresponding to a particular PN code hypothesis, and with each row of an array corresponding to a particular Doppler shift hypothesis. Each array represents the combined results derived from multiple sets of samples. 
     A method of operation of a matched filter in accordance with the subject invention is illustrated in  FIGS. 11A–11C . In step  1000 , a frame of samples is stored. In the matched filter of  FIG. 5 , the frame of samples is stored in RAM  400 . 
     In step  1001 , a PN code hypothesis is selected for testing and stored in a circular shift register. In the matched filter of  FIG. 5 , the PN code hypothesis is stored in PN code register  415 . 
     In step  1002 , a subframe of the frame of samples stored in step  1000  is selected. 
     In step  1003 , a Doppler shift hypothesis is selected for testing. In the matched filter of  FIG. 5 , this step is implicitly performed by Doppler shift generator  401  which successively generates Doppler shift hypothesis responsive to constraints or bounds imposed by a user. In one implementation example, the Doppler shift hypotheses range from ±62,000 Hz. 
     In step  1004 , the subframe selected in step  1002  is corrected for the Doppler shift hypothesis selected in step  1003 . In the matched filter of  FIG. 5 , this step is performed by Doppler shift correction circuitry  403 . In one implementation example, this step is performed as described in U.S. Ser. No. 09/145,055, filed Sep. 1, 1998, now U.S. Pat. No. 6,044,105 previously incorporated herein by reference. 
     In step  1005 , the corrected data from step  1004  is stored, In the matched filter of  FIG. 5 , the corrected data is stored in sample register  406 . 
     In step  1006 , a code phase hypothesis is selected for testing. In the matched filter of  FIG. 5 , this step is implicit in the operation of PN code register  415  which successively and circularly shifts through each of the possible code phase hypotheses in a PN code repetition period, which, in one implementation, comprises 2046 half-chip increments. 
     In step  1007 , the PN code hypothesis selected and stored in step  1001  is circularly shifted by an amount derived from the code phase hypothesis selected in step  1006 . In the matched filter of  FIG. 5 , the selected code phase hypothesis ranges from 0 to 2045 half-chip increments, and step  1007  is implemented by circularly shifting the PN code hypothesis by the number of half-chip increments comprising the selected code phase hypothesis. 
     In step  1008 , the product of the shifted PN code from step  1007 , and the corrected subframe of samples from step  1004  is obtained. In one implementation, this step comprises forming the I and Q component sum of products, SI and SQ. In one implementation example, SI and SQ are derived in accordance with equations (3) and (4), presented earlier. In the matched filter of  FIG. 5 , this step is performed by sum of products circuitry  407 . 
     In step  1009 , the square root of the sum of the squares of SI and SQ from step  1008  is determined in accordance with equation (5), presented earlier. In the matched filter of  FIG. 5 , this step is performed by sqrt of sum of products circuitry  410 . 
     In step  1010 , the value determined in step  1009  is added to any similar values derived from the same hypotheses for previous subframes of the frame which is the subject of step  1000 , and the combined result in then stored. In the matched filter of  FIG. 5 , this step is performed by the adder  411  in combination with RAM  413 , with the combined values being maintained in RAM  413 . 
     In step  1011 , a determination is made whether there are any further code phase hypotheses which are to be tested for the selected PN code and Doppler shift hypotheses. If so, a jump is made to step  1006  and the process beginning at this point repeated for the new code phase hypothesis. If not, step  1012  is performed. In the matched filter of  FIG. 5 , this step is implicit in the operation of PN code register  414 , which successively shifts through the 2046 code phase hypotheses to be tested for a given PN code and Doppler shift hypothesis. 
     In step  1012 , a determination is made whether there are any further Doppler shift hypotheses which are to be tested for the selected PN code hypothesis. If so, a jump is made to step  1003 , and the process beginning at this point repeated for the new Doppler shift hypothesis. If not, step  1013  is performed. In the matched filter of  FIG. 5 , this step is implicit in the operation of Doppler shift generator  401 , which cycles through a plurality of Doppler shift hypotheses for a given PN code hypothesis. In one implementation example, the Doppler shift hypotheses tested for a given PN code hypothesis range from ±62,000 Hz. 
     In step  1013 , a determination is made whether there are any further subframes to be analyzed of the frame which is the subject of step  1000 . If so, a jump is made to step  1002  at and the process beginning at this point repeats itself using the new subframe. If not, the process terminates. In one implementation, the method illustrated in  FIGS. 11A–11C  is then repeated for each PN code hypothesis which is to be tested. In one implementation example, the coordination of this task is performed by GPS processor  303 . 
     Next, the discussion turns to a suitable method for combining the data. An embodiment of method of updating the cumulative correlation data with the correlation data derived from a new segment in accordance with the subject invention is illustrated in  FIG. 9 . In step  800 , a portion of the cumulative correlation data which is to be updated is obtained. In one implementation, this portion is the cumulative correlation data which corresponds to a particular combination of PN code and Doppler shift hypotheses. 
     In step  801 , the corresponding portion of the incremental correlation data, that is, the correlation data derived from a new segment, is obtained. In one implementation, this portion is the incremental correlation data which corresponds to a particular combination of PN code and Doppler shift hypotheses. 
     In step  802 , the code phase difference, ΔCP, between the two portions is determined. In one implementation, this value is derived from the Doppler shift hypothesis corresponding to the two portions, and the time difference between the start of a first segment of samples and a second segment of samples. In one implementation example, ΔCP is determined in accordance with equations (1) and (2) above. 
     In step  803 , the portion of the incremental correlation data obtained in step  801  is adjusted responsive to the code phase difference between the two portions determined in step  802 . In one implementation, this step comprises circularly shifting a row of the incremental data by a value derived from the code phase difference determined in step  802 . In one implementation example, the value ΔCP is computed using equations (1) and (2) above, and the shift value, SHIFT, is determined from the following equation:
 
SHIFT=−Δ CP  modulo 2046  (6)
 
     In step  804 , the portion of the cumulative correlation data obtained in step  800  is updated with the adjusted data determined in step  804 . In one implementation, this step comprises adding a row of circularly shifted incremental data element by element to a corresponding row of cumulative data. 
     In step  805 , a determination is made whether there are any additional portions of cumulative data to update. If so, the process repeats beginning with step  800  for another portion of the cumulative data which is to be updated. If not, the process ends as indicated by identifying numeral  806 . 
     A second embodiment of a method of updating cumulative correlation data with incremental correlation data is illustrated in  FIGS. 8A–8B . In this embodiment, it is assumed that each of the cumulative data and the incremental data are of the form shown in  FIG. 6 . 
     Again, the process is iterative, and each cycle of the process begins at step  700 , in which a PN code hypothesis is selected. Then, in step  701 , the cumulative correlation array for the selected PN code hypothesis is obtained, and in step  702 , the incremental correlation array for the selected PN code hypothesis is obtained. 
     In step  703 , a Doppler shift hypothesis is selected, and in step  704 , the code phase difference, ΔCP, for the selected Doppler shift hypothesis is determined. In one implementation, this value is determined in accordance with equations (1) and (2) above. 
     In step  705 , the corresponding row of the incremental array obtained in step  702  is circularly shifted by an amount derived from the code phase difference determined in step  704 . In one implementation, this value is determined in accordance with equation (6) above. 
     Then, in step  706 , the shifted row from step  705  is added element by element to the corresponding row in the cumulative array. 
     In step  707 , a determination is made whether there are additional Doppler shift hypotheses to be updated. If so, a jump is made to step  703  whereupon the process beginning at this step is repeated for one of the additional Doppler shift hypotheses to be updated. If not, step  708  is performed. In step  708 , a determination is made whether there are additional PN code hypotheses to be updated. If so, a jump is made to step  700  whereupon the process beginning at this step is repeated for one of the additional PN code hypotheses to be updated, the process terminates, as indicated by identifying numeral  709 . 
     IMPLEMENTATION EXAMPLE 
     A matched filter chip code-named “Magna” which combines the functionality of the sampling circuitry  308 , the timing circuitry  307 , offset measurement circuit  305 , and the matched filter  310  of  FIG. 4  has been developed by the assignee of the subject application (Conexant Systems, Inc. of Newport Beach, Calif.). A processor chip which embodies the functionality of the GPS processor  303  of  FIG. 4  code-named “Scorpio”, Part No. 11577-11, is available from the assignee of the subject application. In one implementation, the processor has additional GPS-specific circuits, such as tracking channels for continuously tracking a number of GPS satellite signals. Typically, the processor includes at least an embedded microprocessor with an external bus. In one configuration, the processor views the matched filter chip as a memory mapped peripheral. It issues commands to the matched filter chip, and retrieves results after it has completed processing for a given set of commands. An RF receiver chip which embodies the functionality of the GPS radio receiver  300  of  FIG. 4  code-named “Gemini/Pices Monopack”, Part No. R6732-13, is available from the assignee of the subject application. Additional details regarding this implementation example are available in U.S. Ser. No. 09/145,055, filed in Sep. 1, 1998, now U.S. Pat. No. 6,044,105 previously incorporated herein by reference. 
     The foregoing embodiments, implementations, and implementation examples improve the signal to noise ratio (SNR) of a correlation array through repeated application of a matched filtering operation. The approach described has a number of advantages, which are as follows: 
     1. The embodiments, implementations, and examples are adaptable to any time base. This means that they can be applied to GPS chip sets, cellular and PCS chip sets, and standard microprocessors. 
     2. The embodiments, implementations, and implementation examples allow for the combination of non-uniform receiver sample capture lengths. This is critical for integrated applications such as cellular and PCS, in which it is desirable to receive GPS when the phone is not transmitting. Since the available idle slots will have different durations in the various phone standards, adaptability of time intervals is important. 
     3. The embodiments, implementations, and implementation examples allow arbitrary offsets in the start of the data capture times. This again is most important for phone applications, but it is also important for the basic GPS application. In the GPS application, a flexible start time capability allows the same capture to be used with any satellite in the received samples, regardless of their relative code phases. 
     4. Multiple data capture combining allows SNR to be built up incrementally. After each increment is added, the arrays can be tested for signal detection, and processing for successful acquisitions curtailed. 
     5. Processor RAM and throughput is minimized. This is important for phone applications, in which the baseband device&#39;s digital signal processor (DSP) and protocol stack processors may be busy and RAM limited. For high SNR cases, the entire required RAM for a given PN code is located on or within the matched filter. When data combining is required to improve SNR, only those satellite that have not been detected need be processed. Further, it is often possible in phone systems to reduce the size of the array to less than 2046 elements per PN code and Doppler shift hypothesis, and to a few Doppler shift hypotheses, using information that is inherent in mobile phone network operation. 
     6. While the discussion here has focused on GPS, the invention can be applied to any system using correlation methods for detecting signals. This would include most spread spectrum systems, and general signal detectors. 
     7. Although embodiments and implementations have been described which utilize a GPS processor to perform various functions, it should be understood that embodiments are possible in which a generic processor is used to perform these functions. For purposes of this disclosure, a generic processor is defined to mean any device, including a computer, DSP, baseband processor, microprocessor, or microcomputer, which is capable of executing a discrete series of instructions stored in a memory accessible by the processor. It should also be understood that embodiments are possible in which analog circuitry is used to perform these functions. 
     While various embodiments of the application have been described, it will be apparent to those of ordinary skill in the art that many more embodiments and implementations are possible that are within the scope of the subject invention. Accordingly, the invention is not to be restricted except in light of the appended claims and their equivalents.