Abstract:
An analog input circuit may include a pair of differential transconductance input stages having input nodes connected in parallel and which are fed the analog input signal. One of the differential transconductance stages may have common mode compatibility toward the supply node at the highest potential, and the other stage may have common mode compatibility toward the supply node at the lowest potential. Furthermore, differential output currents of the transconductance input stages may be summed differentially on first and second input nodes of a differential converter stage, which converts the differential current signals to an amplified differential voltage output signal.

Description:
FIELD OF THE INVENTION 
     The present invention relates to the field of integrated circuits, and, more particularly, to an input circuit with common mode compatibility with respect to its supply nodes. 
     BACKGROUND OF THE INVENTION 
     In various applications relating to analog signal processing, the common mode of the input signal may vary from the positive to the negative supply voltage such as, for example, with standard low voltage differential signals (LVDS) having variable voltages from 0 to 2.4V. For these applications, the input stages of the interface circuits should be able to accept a signal with such variations without significantly degrading the performance of the amplifier. 
     A typical application in which the circuit of the invention may be used is illustrated in FIGS. 1A and 1B. In the illustrated example, a digital signal, transmitted on an optical fiber, is received by a first integrated circuit IC 1  installed on a printed circuit board. After having been processed, the signal is sent from the first integrated circuit IC 1  by a transmission block Transmit Tx through the pins and the connecting metal tracks PCB tracks to a second integrated circuit IC 2 . The integrated circuit IC 2  is installed on the same board and receives the signal by way of an input circuit thereof which amplifies the received analog signal to make it available at an appropriate level to logic circuitry CMOS logic. 
     While transmission of signals through the conducting tracks of the printed circuit board is commonly done in a differential mode according to standard LVDS, the input stage of the integrated circuit IC 2  that receives the signal processed by the interfacing circuit IC 1  may be single ended, as shown in FIG. 1A, or may have a differential output, as shown in FIG.  1 B. If the input interfacing circuit IC 1  is single ended, the output signal will have a certain average value V avg . If the output is differential, the output signal will be centered on a certain common mode voltage V cm . 
     Other circuits are commonly connected in cascade to the input interfacing circuit IC 1 , the first of which may be an analog gain circuit G. The gain circuit G is followed by a digital buffer stage which, in practice, may even be represented by a single circuit (e.g., an inverter) having a certain threshold voltage for discriminating zeroes and ones. 
     It will be appreciated that, for a correct interpretation of the signals that are fed to the block G, it is important that the amplitude of the signal corresponding to a logic value 1 remains above the threshold of discrimination. Further, the amplitude of the signal corresponding to a zero logic value should also remain below the threshold of discrimination for both single ended and differential signals. 
     FIGS. 2A and 2B are diagrams of the signals output by the input interfacing circuit for the case of a single ended or a differential output circuit when the input signal is an analog signal corresponding to a, certain sequence of ones and zeroes. In practice, the output signal V out  (in the case of single ended output) or V out+  minus V out−  (in the case of differential output) will have a shape more or less rounded and with a voltage swing (i.e., the difference between the maximum and the minimum value) typically between 250-400 mV, as illustratively shown. 
     As also shown in the illustrated examples, in the case of a differential output there will be a certain common mode voltage V cm , while in the case of single ended output there will be an average voltage V avg , both corresponding to the average output voltage following a long sequence of alternated zeroes and ones (FIG.  2 A). The switching threshold of the input stage G of the digital circuitry (FIGS. 1A and 1B) will be fixed to a value to correspond to the average value V avg  or to the common mode value V cm . Therefore, a fundamental prerequisite of the input circuit is that the output signal be centered with respect to these values (V avg  or V cm ), as the threshold of discrimination of the input digital value is fixed as a function thereof. 
     In this way, the level of the output signal may be shifted with respect to the level of the input signal without loosing data because the output signal is kept centered independently from the succession of values of the input signal, as shown in the examples of FIG.  2 B and at (a) in FIG.  3 . If, because of the characteristics of the input circuit, the output signal is not kept centered with respect to its average value or its common mode value, as shown at (c) in FIG. 3, it may happen that a digital value 1 is erroneously read as a zero by the digital circuit connected in cascade to the input circuit (or vice-versa). This may happen because the signal may remain below (or above) the switching threshold if a sufficiently long sequence of zeroes (or ones) determines a shift of the working point of the output node. This shift is either toward ground in the case of a long sequence of zeroes, as shown at (c) in FIG. 3, or toward the supply voltage in the case of a long sequence of ones. 
     A known prior art design approach for the input circuit is illustrated in FIG. 4 which utilizes two operational amplifiers (op-amps), one of which has a common mode compatibility toward the supply voltage (V dd ) and the other toward ground (GND). The voltage outputs are both single ended and connected in common to provide for a single ended output of the entire circuit. 
     The design illustrated in FIG. 4 has certain drawbacks. First, the output is single ended and is obtained by short circuiting the outputs of the two distinct operational amplifiers, which may conflict with each other. This may slow down the system because the circuit has to work even when the two parts are in conflict. This happens because of an unbalanced common mode, either toward the supply voltage (V dd ) or toward ground (GND), which causes one of the two pairs of input transistors of the two distinct operational amplifiers to start turning off. 
     Furthermore, the output node V out  is a high impedance node having a high gain and, as a result, is relatively slow. The high impedance implies that in the presence of certain input sequences there is a likelihood of losing data in the case illustrated at (c) in FIG.  3 . In fact, in an integrated circuit a high impedance output node has a parasitic series capacitance toward the substrate creating a pole at a low frequency. This practically integrates the output signal and centers it on a short-term average value, producing the result shown at (c) in FIG.  3 . 
     Yet another drawback of the above-described prior art design is that two reference voltages are needed, namely V BP2  and V BN2 . The generation of these voltages may require further silicon area for implementation. 
     SUMMARY OF THE INVENTION 
     It is therefore an object of the invention to provide an analog input circuit which provides full input common mode compatibility toward supplies rails and provides for a low impedance output, thus reducing the risk of losing data, and with a circuit configuration that is relatively simple and integrable in a reduced silicon area with respect to prior art circuits. 
     In accordance with the invention, an analog input circuit may include a pair of differential transconductance input stages having input nodes connected in parallel and which are fed the analog input signal. One of the differential transconductance stages may have common mode compatibility toward the supply node at the highest potential, and the other stage may have common mode compatibility toward the supply node at the lowest potential. Furthermore, differential output currents of the transconductance input stages may be summed differentially on first and second input nodes of a differential converter stage, which converts the differential current signals to an amplified differential voltage output signal. 
     The analog input circuit in accordance with the invention has a relatively low output impedance which provides high speed, as well as enhanced stability of the working point of the circuit. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The present invention and the particular features and advantages thereof will be further understood with reference to the following detailed description of an embodiment thereof and with reference to the attached drawings, in which: 
     FIGS. 1A and 1B are schematic block diagrams of printed circuit boards including analog input circuits in accordance with the prior art; 
     FIGS. 2A,  2 B, and  3  are signal diagrams illustrating certain operating conditions of the analog input circuits of FIGS. 1A and 1B. 
     FIG. 4 is a schematic circuit diagram of a prior art analog input circuit including two operational amplifiers connected in parallel; 
     FIG. 5 is a schematic block diagram of an analog input circuit in accordance with the present invention; 
     FIG. 6 is a schematic block diagram of an alternate embodiment of the analog input circuit of FIG. 5; 
     FIG. 7 is a schematic circuit diagram illustrating in greater detail the analog input circuit of FIG. 6; 
     FIG. 8 is a schematic circuit diagram of the analog input circuit of FIG. 7 with relevant signal analysis reference data; 
     FIG. 9 is a schematic circuit diagram of the analog input circuit of FIG. 7 with common mode analysis reference data; and 
     FIG. 10 is a schematic block diagram of the analog input circuit of FIG. 7 with differential mode analysis reference data. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Referring now to FIG. 5, the analog input circuit of the present invention is, generally speaking, based on the use of two transconductance differential input stages g m1  and g m2  whose inputs are connected in parallel. The differential stage g m1  is compatible toward the supply voltage, and the differential stage g m2  is compatible toward ground. The differential output currents I diff1  and I diff2  are summed, in differential mode, and the resulting differential current signal I diff  is converted to a differential voltage signal on the output nodes of the circuit, V out+  and V out− . The circuit illustrated in FIG. 5 may be implemented by using two operational amplifiers OA 1  and OA 2 , in which one of the operational amplifiers includes an output I to V converting stage, as illustrated with a dashed box in FIG.  6 . 
     Referring more particularly to FIG. 7, an exemplary circuit implementation of the circuit of FIG. 6 is illustrated in which the operational amplifiers OA 1  and OA 2  are connected in parallel. As will be described further below, the operational amplifier OA 1  is compatible toward ground potential GND if the process of fabrication of the integrated circuit produces turn on threshold voltages for the P-channel and N-channel transistors of the differential input pair M 3  and M 4  such that V thN &lt;V thP . Similarly, OA 2  is compatible toward the supply voltage V dd  because of a proper sizing of the components thereof, as will also be discussed below. 
     The functioning of the circuit will be further understood with reference to FIGS. 8,  9  and  10 , which show data and parameters relative to the differential input circuit, the common mode analysis, and the differential mode analysis, respectively. The voltage of the differential input signal, 
     
       
           V   in+   −V   in− =2 ΔV   in ,  
       
     
     is applied in parallel to the inputs of the two operational amplifiers OA 1  and OA 2 . In the operational amplifier OA 1 , the resulting differential currents ΔI 1  and −ΔI 1  are output by two current mirrors, CM_ 1  and CM_ 2 , respectively, with a certain mirror ratio K. The differential output voltage, 
     
       
           V   out+   −V   out− =2 ΔV   out ,  
       
     
     is given by: 
     
       
           V   2-R   −V   1-R =2 R ( KΔI   1   +ΔI   2 ).  
       
     
     With particular reference to FIG. 9, the output common mode voltage is given by: 
     
       
           V   out cm   =V   dd   −R ( I   G2   +K I   G1 ),  
       
     
     and may be predetermined by choosing R, K, I G1  and I G2 . 
     The common mode compatibility toward ground as well as toward the supply voltage will now be further described. Starting the analysis from the operational amplifier OA 1  (FIG.  8 ), the gate-source voltage of the MOS transistor M 1  (and similarly of M 2 ) is given by: 
     
       
           V   GS1   =V   thN   +V   od1 ,  
       
     
     where the threshold voltage for an N channel MOS is defined as V thN &gt;0, and V od1  is the overdrive voltage of M 1  which produces, for the chosen aspect ratio W 1 /L 1 , the current I G1  when M 1  saturates. 
     For saturation, it is necessary that V DS1  be such that: 
     
       
           V   GS1   −V   thN   =V   od1 ( I   G1 )≦ V   DS1  (with  V   DS1 &gt;0), and  
       
     
     
       
         − V   thN   ≦V   DS1   −V   GS1 ;  
       
     
     i.e., if: 
     −V thN ≦V DG1 . 
     Of course, the diode connection of M 1  ensures that such a condition be verified. The value of the overdrive voltage V od1  is given by the well known relation: 
     
       
           I   D   =K ( W/L )( V   GS   −V   th ) 2   =K ( W/L ) V   2   od ,  
       
     
     that correlates the overdrive voltage to the current delivered (for PMOS and NMOS transistors). The sum of the (common mode) voltages on M 3  and M 1  is given by: 
     
       
           V   G1   +V   DS3   +V   DS1   =V   dd ;  
       
     
     
       
           V   G1   +V   GS3   +V   GD3 +( V   thN   +V   od1 )= V   dd ; and  
       
     
     
       
           V   G1 +( V   thP   +V   od3 )+ V   GD3 +( V   thN   +V   od1 )= V   dd ,  
       
     
     where V thP  and V od3  are the threshold and the overdrive voltages of the MOS transistor M 3 , for which the same considerations made for the transistor M 1  hold. The threshold voltage for a PMOS transistor is defined as V thP &lt;0, and also V od3  is negative. 
     The conditions under which the operational amplifier OA 1  is compatible toward ground potential GND will now be discussed, starting from a saturated working condition. To this end, let us suppose that the input common mode voltage V in cm  decreases (FIGS.  8  and  9 ). The current I G1  remains constant, thus the voltages V GS1  and V GS3  remain constant while the voltages V DS3  and V GD3  decrease and the voltage V G1  increases. This may continue as long as M 3  and M 1  reach saturation, that is as long as V GD3 ≧V thP  (i.e., considering the negative sign of V thP , |V GD3 |≦|V thP |, V GD3  being negative) and consequently V DS3 ≧−V od3  (that is V DS3 ≧|V od3 |). 
     Should it happen that, while V in cm  continues to decrease, the voltage V GD3  diminishes below the negative value of the threshold voltage V thP , then M 3  would exit from saturation and enter in the linear zone of its characteristics. The current I G1  remains constant, but the voltage V GS3  increases, the voltage V GD3  diminishes below V thP  (i.e., V GD3 &lt;−|V thP | and is positive), and the sum voltage V DS3  diminishes, but less rapidly than when the transistor M 3  was saturated. In other words, for a current I D3 =I G1 =a constant, the working point shifts along characteristic curves corresponding to increasing values of V GS . The voltage V G1  increases, but by a small amount. 
     The transistor M 1  continues to work in the saturation region. Obviously this behavior is undesirable in this case because an amplifier operates correctly only if both MOS transistors M 3  and M 4  are saturated. The limit condition for saturating M 3  and M 4 , considering the limit of the saturating condition (for which V GD3 =V thP ), is: 
     
       
           V   in cm   =V   GD3   +V   DS1   =V   thP   +V   thN   +V   od1 ≦0.  
       
     
     Thus, for compatibility of the common mode with respect to the ground potential (V in cm =0), it is desired that the following condition be verified: 
     
       
           V   thP   +V   thN   +V   od1 ≦0, and  
       
     
     
       
         | V   thP   |≧V   thN   +V   od1 ,  
       
     
     (because V thP  has been defined to be negative). 
     By contrast, the operational amplifier OA 1  is not compatible with respect to the supply voltage V dd . In fact, V in cm  may increase only as far as the voltage on the current generator G 1  will permit its functioning as a current generator. Beyond that, I G1  starts decreasing and the amplifier progressively turns off, while the MOS transistors M 3  and M 1  continue to be in saturation, but with gradually decreasing currents. 
     This may be observed by considering that the generator G 1  is made by P-channel MOS transistors, for which the same considerations made above hold. That is, the condition V G1 ≧|V odG1 | should be verified for the MOS transistors to operate in the saturation region. When V G1  diminishes the overdrive voltage diminishes, and as a consequence the delivered current also diminishes until the MOS transistors turn off. It is worth noting that the turn off is not abrupt but gradual. This represents a particular advantage in many applications. Of course, similar considerations are valid also for the transistors M 4  and M 2 . 
     Compatibility with respect to the supply voltage is provided by the second operational amplifier OA 2 . Similar to what has been discussed with reference to the operation amplifier OA 1 , the voltage V DG6  may become negative, but the absolute value thereof should not surpass V thN , thus it is necessary that: 
     
       
           V   thN   ≧R ·( I   G2   +K·I   G1 ).  
       
     
     If this relation is verified, the transistor M 6  correctly operates in the saturation region even when V out cm =V dd , otherwise it would operate in the linear functioning region. Similar considerations are also valid for the transistor M 7 . 
     Summarizing the above considerations, compatibility toward the supply voltage is provided by OA 2  by virtue of an adequate sizing of the MOS transistors and of resistors R. Compatibility toward ground potential, ensured by OA 1 , requires that the overdrive voltage of the N-channel CMOS transistors that form the mirrors CM_ 1  and CM_ 2  be sufficiently small, and it is desirable that |V thP | be greater than V thN . 
     Among the numerous advantages of the analog input circuit of this invention, it is particularly worth noting that this circuit has a relatively simple design, is easily realized without other ancillary circuitry, it has low output impedance, and it requires relatively less area for integration.