Abstract:
For use with a power converter couplable to a source of electrical energy, the converter having a power switch that conducts intermittently to transfer energy from the source to an inductive element, and a freewheeling diode that alternately conducts with the power switch to transfer energy to an output of the converter, an active clamp and a method of operating the active clamp. In one embodiment, the active clamp includes: (1) an inductor, coupled in series with the freewheeling diode, and (2) a series-coupled capacitor and clamping switch, coupled in parallel with the inductor, that cooperate therewith to mitigate adverse effects of reverse recovery currents associated with the freewheeling diode and enable substantially zero voltage switching of the power and clamping switches.

Description:
TECHNICAL FIELD OF THE INVENTION 
     The present invention is directed, in general, to power conversion and, more specifically, to an active clamp for a power converter, a method of operating the active clamp and a power converter employing the active clamp or the method. 
     BACKGROUND OF THE INVENTION 
     A power converter is a power processing circuit that converts an input voltage or current waveform into a specified output voltage or current waveform. A switched-mode power converter is a frequently employed power converter that converts an input voltage waveform into a specified output voltage waveform. A buck converter is one example of a switched-mode power converter that is typically employed in applications wherein a stable, regulated voltage is desired at the output of the power converter. 
     A non-isolated buck converter generally includes a power switch couplable to a source of input voltage. The power switch intermittently switches to provide an output voltage to a load couplable to an output of the buck converter. A controller regulates the output voltage by varying a duty cycle of the power switch. Depending on the duty cycle of the power switch, the output voltage may be regulated to any desired voltage between zero and the input voltage. 
     The controller typically switches the power switch at a high switching frequency, such as one beyond the audible range, to reduce the size and weight of inductive components employed and, therefore, to reduce the cost, as well as the size and weight, of the buck converter. Conventional buck converters, therefore, typically include a low pass output filter having a filter inductor and a filter capacitor. The comer frequency of the output filter may be set sufficiently lower than the switching frequency of the power switch to minimize the output ripple. 
     Since the power switch is coupled in series with the filter inductor, turning off the power switch may result in a high voltage thereacross unless an alternative path is provided for the inductor current. A freewheeling diode may, therefore, be coupled between common and a node between the power switch and the filter inductor to provide a path for the inductor current while the power switch is off. During a conduction interval of the power switch, the freewheeling diode is reversed biased. Then, during a non-conduction interval of the power switch, the inductor current flows through the freewheeling diode, transferring some of its stored energy to the load. The buck converter, like other switched-mode power converters, preferably includes at least two semiconductor switches, the power switch and the freewheeling diode. 
     Analogous to other types of power converters (e.g., a boost converter), the buck converter is subject to inefficiencies that impair its overall performance. More specifically, the power switch and freewheeling diode may be subject to conduction losses that reduce the efficiency of the converter. Additionally, the power switch [e.g., a metal-oxide semiconductor field-effect transistor (MOSFET)] is subject to switching losses that occur, in part, when a charge built-up in a parasitic capacitance of the power switch is dissipated during turn-on. Furthermore, the freewheeling diode may also be subject to a reverse recovery condition (when the power switch is turned on) that induces a substantial current spike through both the power switch and the freewheeling diode. The losses associated with the power switch and the freewheeling diode increase linearly as the switching frequency of the converter is increased. Therefore, minimizing the reverse recovery and switching losses associated with the freewheeling diode and power switch will improve the overall efficiency of the converter. 
     Accordingly, what is needed in the art is an active clamp, employable with a variety of power converter topologies, that reduces the losses associated with the reverse recovery condition and further reduces the switching losses associated with the power switch of a power converter. 
     SUMMARY OF THE INVENTION 
     To address the above-discussed deficiencies of the prior art, the present invention provides, for use with a power converter couplable to a source of electrical energy, the converter having a power switch that conducts intermittently to transfer energy from the source to an inductive element, and a freewheeling diode that alternately conducts with the power switch to transfer energy to an output of the converter, an active clamp and a method of operating the active clamp. In one embodiment, the active clamp includes: (1) an inductor, coupled in series with the freewheeling diode, and (2) a series-coupled capacitor and clamping switch, coupled in parallel with the inductor, that cooperate therewith to mitigate adverse effects of reverse recovery currents associated with the freewheeling diode and enable substantially zero voltage switching of the power and clamping switches. 
     The present invention, in one aspect, introduces the broad concept of an active clamp employable with switched-mode power converter topologies having a freewheeling diode subject to reverse recovery currents. The active clamp is capable of reducing the reverse recovery losses associated with the freewheeling diode and is further capable of reducing the switching losses associated with the power switch of the power converter. 
     In one embodiment of the present invention, the clamping switch conducts to couple the capacitor across the inductor, thereby enabling the capacitor to discharge through the inductor. Consequently, the amount of energy stored in the capacitor is sufficient to enable substantially zero voltage switching of the power and clamping switches. 
     In an embodiment to be illustrated and described, the clamping switch is a metal oxide semiconductor field-effect transistor (MOSFET). Those skilled in the pertinent art will understand, however, that the present invention fully encompasses all controllable switches, whether conventional or later-developed. In a related embodiment, the freewheeling diode and a body diode of the clamping switch are similarly oriented. Of course, an external diode may be employed as required. 
     In one embodiment of the present invention, the converter is selected from the group consisting of a non-isolated boost converter, a non-isolated buck converter, a non-isolated buck-boost converter, a non-isolated capacitive-coupled buck-boost converter, a non-isolated Sepic converter, and a non-isolated Zeta converter. Those skilled in the pertinent art understand, however, that other converter topologies are well within the broad scope of the present invention. 
     In one embodiment of the present invention, the converter further includes a controller coupled to the power and clamping switches. The controller controls conduction intervals of the power and clamping switches. In an embodiment to be illustrated and described, the controller monitors the output voltage of the converter and controls the power and clamping switches in response thereto. Of course, those skilled in the pertinent art understand that other control points within the converter, or power supply as a whole, are within the broad scope of the present invention. 
     The foregoing has outlined, rather broadly, preferred and alternative features of the present invention so that those skilled in the art may better understand the detailed description of the invention that follows. Additional features of the invention will be described hereinafter that form the subject of the claims of the invention. Those skilled in the art should appreciate that they can readily use the disclosed conception and specific embodiment as a basis for designing or modifying other structures for carrying out the same purposes of the present invention. Those skilled in the art should also realize that such equivalent constructions do not depart from the spirit and scope of the invention in its broadest form. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     For a more complete understanding of the present invention, reference is now made to the following descriptions taken in conjunction with the accompanying drawings, in which: 
     FIG. 1 illustrates a schematic diagram of an embodiment of a non-isolated buck converter constructed in accordance with the principles of the present invention; 
     FIG. 2A through 2H illustrate graphical representations of a plurality of current and voltage waveforms of the non-isolated buck converter of FIG. 1; 
     FIG. 3 illustrates a schematic diagram of an embodiment of a non-isolated boost converter constructed in accordance with the principles of the present invention; 
     FIG. 4 illustrates a schematic diagram of an embodiment of a non-isolated buck-boost converter constructed in accordance with the principles of the present invention; 
     FIG. 5 illustrates a schematic diagram of another embodiment of a non-isolated buck-boost converter constructed in accordance with the principles of the present invention; 
     FIG. 6 illustrates a schematic diagram of an embodiment of a non-isolated Sepic converter constructed in accordance with the principles of the present invention; 
     FIG. 7 illustrates a schematic diagram of an embodiment of a non-isolated capacitive-coupled buck-boost converter constructed in accordance with the principles of the present invention; and 
     FIG. 8 illustrates a schematic diagram of an embodiment of a non-isolated Zeta converter constructed in accordance with the principles of the present invention. 
    
    
     DETAILED DESCRIPTION 
     Referring initially to FIG. 1, illustrated is a schematic diagram of an embodiment of a non-isolated buck converter  100  constructed in accordance with the principles of the present invention. The buck converter  100  has an input couplable to a source of electrical power  105  having an input voltage V IN  and an output couplable to a load  190 . The buck converter  100  includes a power switch Q 1  coupled to the input. In the illustrated embodiment, the power switch Q 1  is a metal oxide semiconductor field-effect transistor (MOSFET). Of course, other controllable switches, such as bipolar junction transistors (BJTs), insulated gate bipolar transistors (IGBTs) and gate turn-off thyristors (GTOs) are well within the broad scope of the present invention. 
     The buck converter  100  further includes an output filter  180  having a filter inductor (inductive element) L F  and a filter capacitor C F . In the illustrated embodiment, the output filter  180  is a low pass filter having a comer frequency set sufficiently lower than the switching frequency of the power switch Q 1  to minimize ripple in an output voltage V OUT . The buck converter  100  further includes a freewheeling diode D 1  coupled to a node  120  between the power switch Q 1  and the filter inductor L F . 
     The buck converter  100  further includes an active clamp  130  coupled to the freewheeling diode D 1 . The active clamp  130  includes an inductor (clamping inductor L C ) coupled in series with the freewheeling diode D 1 . The active clamp  130  further includes a series-coupled capacitor (clamping capacitor C C ) and clamping switch Q 2  coupled in parallel with the clamping inductor L C . In the illustrated embodiment, the clamping switch Q 2  is a MOSFET having a body diode D 2 . The clamping switch Q 2  may be positioned such that the body diode D 2  is similarly oriented with respect to the freewheeling diode D 1 . While the body diode D 2  is integral to the clamping switch Q 2  and is explicitly illustrated only to show its orientation, an external diode may be employed as required to supplement the body diode D 2 . Those skilled in the pertinent art will realize that, while the power switch Q 1  and the clamping switch Q 2  are illustrated as MOSFETs, the use of any conventional or later-developed controllable switch is well within the broad scope of the present invention. 
     The buck converter  100  still further includes a controller  150 , coupled to the power and clamping switches Q 1 , Q 2 , that controls conduction intervals of the power and clamping switches Q 1 , Q 2 . In the illustrated embodiment, the controller  150  monitors the output voltage V OUT , and controls the power and clamping switches Q 1 , Q 2  to regulate the output voltage V OUT  of the buck converter  100 . Of course, the controller  150  may monitor other control points within the buck converter  100  as desired. 
     Turning now to FIGS. 2A through 2H, illustrated are graphical representations of a plurality of current and voltage waveforms of the buck converter  100  of FIG.  1 . More specifically, FIG. 2A illustrates a gate voltage V GQ1  of the power switch Q 1 . As illustrated, the power switch Q 1  is conducting during a primary interval D (t 4 −t 0 ) and is non-conducting during an complementary interval 1−D (t 0 −t 4 ). FIG. 2B illustrates a drain current I Q1  through the power switch Q 1 . FIG. 2C illustrates a freewheeling current I D1  through the freewheeling diode D 1 . FIG. 2D illustrates a clamping inductor current I LC  through the clamping inductor L C . FIG. 2E illustrates a body diode current I D2  through the body diode D 2  of the clamping switch Q 2 . FIG. 2F illustrates a gate voltage V GQ2  of the clamping switch Q C . FIG. 2G illustrates a drain current I Q2  through the clamping switch Q 2 . FIG. 2H illustrates a drain-source voltage V DSQ1  across the power switch Q 1 . 
     With continuing reference to FIG. 1, the buck converter  100  operates as follows. The power switch Q 1  is conducting during the primary interval D (t 4 −t 0 ; see FIG. 2A) to transfer a portion of the electrical power from the source  105  to the filter inductor L F  and to the load  190 . During the primary interval D, the drain current I Q1  flows through power switch Q 1 , and the filter inductor L F  (see FIG.  2 B). The freewheeling diode D 1  is reverse biased (see FIG. 2C) and there is negligible clamping inductor current flow through the clamping inductor L CLAMP  (see FIG.  2 D). 
     The power switch Q 1  is then turned off at the end of the primary interval D (at t 0 ; see FIG.  2 A). An inductor current I LF  (through the filter inductor L F ) established by the conduction of the power switch Q 1  during the primary interval D initially (at t 0 ) flows to the output through the body diode D 2  of the clamping switch Q 2  (see FIG.  2 E), the clamping capacitor C C  and the freewheeling diode D 1  (see FIG.  2 C). 
     As the inductor current I LF  flows through the clamping capacitor C C , the clamping capacitor C C  begins to charge and a clamping voltage is developed thereacross. Since the clamping capacitor C C  and clamping switch Q 2  are coupled across the clamping inductor L C , the clamping voltage is applied across the clamping inductor L C , causing the clamping inductor current I LC  (which is a portion of the inductor current I LF ) to begin to flow through the clamping inductor L C  (see FIG.  2 D). As the clamping capacitor C C  continues to charge, the body diode current I D2  through the body diode D 2  and clamping capacitor C C  decreases (see FIG. 2E) while the clamping inductor current I LC  increases (see FIG.  2 D). Once the current through the clamping capacitor C C  has decreased to zero, the clamping inductor current I LC  will be equal to the inductor current I LF  until the clamping switch Q 2  is turned on (at t 2 ). The clamping capacitor C C  can now discharge through the clamping switch Q 2  and the clamping inductor L C . 
     The clamping capacitor C C  replenishes its charge at two distinct time intervals (t 0 −t 1  and t 5 −t 6 ) during a switching cycle. First, the clamping inductor current I LC  may flow through the body diode D 2  of the clamping switch Q 2  to charge the clamping capacitor C C  (t 0 −t 1 ; see FIGS. 2D,  2 E). Second, a reverse recovery current from the freewheeling diode D 1  may flow into the clamping inductor L C  to store energy in the clamping inductor L C . The energy stored in the clamping inductor L C  may then be recovered into the clamping capacitor C C  via the body diode D 2  once the freewheeling diode D 1  has completed the reverse recovery period (t 5 −t 6 ). 
     The clamping switch Q 2  is turned on for a short interval (t 2 −t 3 ) at the end of a complementary interval 1−D (see FIG.  2 F). During this interval (t 2 −t 3 ), the clamping voltage (available across the clamping capacitor C C ) is applied across the clamping inductor L C  to cause the clamping inductor current I LC  to ramp up above the inductor current I LF  (see FIG.  2 D). The clamping switch Q 2  is then turned off (at t 3 ). A difference in current between the clamping inductor current I LC  and the inductor current I LF  causes the voltage at the node  120  to rapidly increase until it is substantially equal to the input voltage V IN . The drain source voltage V DSQ1  across the power switch Q 1  decreases rapidly to substantially zero (t 3 −t 4 ; see FIG.  2 H). The power switch Q 1 , may now be turned on with substantially zero volts thereacross to initiate the primary interval D (at t 4 ; see FIG.  2 H). Conventionally, the conduction of the power switch Q 1  would cause the freewheeling diode D 1  to exhibit reverse recovery currents for a short time (during t 4 −t 5 ) as current from the source  105  flows in a reverse direction through the freewheeling diode D 1 . The present invention, in an advantageous embodiment, positions the clamping inductor L C  in series with the freewheeling diode D 1  such that the clamping inductor L C  effectively restricts the reverse recovery current, thus reducing energy losses due to the reverse recovery condition. 
     Turning now to FIG. 3, illustrated is a schematic diagram of an embodiment of a non-isolated boost converter  300  constructed in accordance with the principles of the present invention. The boost converter  300  has an input couplable to a source of electrical power  305  having an input voltage V IN  and an output couplable to a load  390 . The boost converter  300  includes a filter inductor L F  coupled to the input. The boost converter  300  further includes a power switch Q 1  coupled to the filter inductor L F . The boost converter  300  further includes a freewheeling diode D 1  coupled to a node  320  between the filter inductor L F  and the power switch Q 1 . The boost converter  300  further includes a filter capacitor C F  coupled across the output. The filter capacitor C F  is preferably large to maintain an output voltage V OUT  at a substantially constant level. 
     The boost converter  300  further includes an active clamp  330  coupled between the freewheeling diode D 1  and the node  320 . In the illustrated embodiment, the active clamp  330  includes a clamping inductor L C  coupled in series with the freewheeling diode D 1 . The active clamp  330  further includes a series-coupled clamping capacitor C C  and clamping switch Q 2  coupled in parallel with the clamping inductor L C . In the illustrated embodiment, the clamping switch Q 2  is a MOSFET having an integral body diode D 2 . The body diode D 2  is explicitly illustrated to indicate that its orientation is similar to that of the freewheeling diode D 1 . Those skilled in the pertinent art realize that, while the illustrated body diode D 2  is integral to the clamping switch Q 2 , an external diode may be employed as required to supplement the body diode D 2 . Those skilled in the pertinent art also realize that the power switch Q 1  and the clamping switch Q 2  may be any controllable switch, whether conventional or later-developed. 
     The boost converter  300  still further includes a controller  350 , coupled to the power switch Q 1  and the clamping switch Q 2 , that controls conduction intervals of the power and clamping switches Q 1 , Q 2 . In the illustrated embodiment, the controller  350  monitors the output voltage V OUT  and controls the power and clamping switches Q 1 , Q 2  to regulate the output voltage V OUT  of the boost converter  300 . Of course, the controller  350  may monitor other control points within the boost converter  300  (e.g., an input current or an output current) as desired. 
     The boost converter  300  operates as follows. During a primary interval D, the input voltage V IN  supplies energy via the power switch Q 1  to charge the filter inductor L F . A filter inductor current I LF  flows through the filter inductor L F  and the power switch Q 1  . The freewheeling diode D 1  is reverse biased, decoupling the filter capacitor C F  and the load  390  from the source  305 . During the primary interval D, a stored charge in the filter capacitor C F  provides the output voltage V OUT  to the load  390 . 
     The power switch Q 1  then turns off at the end of the primary interval D. A voltage across the filter inductor L F  reverses while the filter inductor current I LF  maintains its direction of flow. The voltage at the node  320  increases, causing the filter inductor current I LF  to initially flow through the clamping capacitor C C  and the body diode D 2  to the output of the boost converter  300 . 
     As the filter inductor current I LF  flows through the clamping capacitor C C , the clamping capacitor C C  begins to charge and a clamping voltage is developed thereacross. Since the clamping capacitor C C  and clamping switch Q 2  are coupled across the clamping inductor L C , the clamping voltage is applied across the clamping inductor L C , causing a portion of the filter inductor current I LF  to begin to flow through the clamping inductor L C . Then, as the clamping capacitor C C  continues to charge, a body diode current I D2  through the body diode D 2  and the clamping capacitor C C  decreases while a clamping inductor current I LC  increases. Once the current through the clamping capacitor C C  has decreased to zero, the clamping inductor current I LC  will be equal to the filter inductor current I LF  until the clamping switch Q 2  is turned on. 
     The clamping switch Q 2  is turned on for a short interval at the end of a complementary interval 1−D. During the complementary interval 1−D, the clamping voltage is applied across the clamping inductor L C  to cause the clamping inductor current I LC  to ramp up above the filter inductor current I LF . The clamping switch Q 2  is then turned off. A difference between the filter inductor current I LF  and the clamping inductor current I LC  causes the voltage at the node  320  to rapidly decrease to common. The power switch Q 1  may now be turned on with substantially zero volts thereacross to initiate the primary interval D. Conventionally, the turn-on of the power switch Q 1  would cause the freewheeling diode D 1  to exhibit a reverse recovery condition for a short time. The present invention, in an advantageous embodiment, positions the clamping inductor L C  in series with the freewheeling diode D 1  such that the clamping inductor current I LC  through the clamping inductor L C  effectively resists the current from the filter capacitor C F  thus reducing energy losses due to reverse recovery. 
     Turning now to FIG. 4, illustrated is a schematic diagram of an embodiment of a non-isolated buck-boost converter  400  constructed in accordance with the principles of the present invention. The buck-boost converter  400  has an input couplable to a source of electrical power  405  having an input voltage V IN  and an output couplable to a load  490 . The buck-boost converter  400  includes a power switch Q 1  coupled to the input. The buck-boost converter  400  further includes an inductor L F  coupled between the power switch Q 1  and common. The buck-boost converter  400  further includes a freewheeling diode D 1  coupled to a node  420  between the power switch Q 1  and the inductor L F . The buck-boost converter  400  further includes a filter capacitor C F  coupled across the output. The filter capacitor C F  is preferably large to maintain an output voltage V OUT  at a substantially constant level. 
     The buck-boost converter  400  further includes an active clamp  430  coupled between the freewheeling diode D 1  and the filter capacitor C F . In the illustrated embodiment, the active clamp  430  includes a clamping inductor L C  coupled in series with the freewheeling diode D 1 . The active clamp  430  further includes a series-coupled clamping capacitor C C  and clamping switch Q 2  coupled in parallel with the clamping inductor L C . In the illustrated embodiment, the power switch Q 1  and the clamping switch Q 2  are illustrated as MOSFETs. Those skilled in the pertinent art realize, of course, that both the power switch Q 1  and the clamping switch Q 2  may be any controllable switch, whether conventional or later-developed. 
     The buck-boost converter  400  still further includes a controller  450  coupled to the power switch Q 1  and the clamping switch Q 2 . In the illustrated embodiment, the controller  450  monitors the output voltage V OUT  and controls the power and clamping switches Q 1 , Q 2  to regulate the output voltage V OUT  of the buck-boost converter  400 . While the illustrated controller  450  monitors the output voltage V OUT , those skilled in the pertinent art realize that the controller  450  may monitor other control points within the buck-boost converter  400  (e.g., an input current or an output current) as desired. 
     The operation of the buck-boost converter  400  is analogous to the operation of the buck converter  100  of FIG.  1  and the boost converter  300  of FIG. 3 and, as a result, the operation thereof will only be briefly described. 
     During a primary interval D, the input voltage V IN  supplies energy via the power switch Q 1  to charge the inductor L F . An inductor current I LF  flows through the power switch Q 1  and the inductor L F . The freewheeling diode D 1  is reverse biased, decoupling the filter capacitor C F  and the load  490  from the source  405 . During the primary interval D, a stored charge in the filter capacitor C F  provides the output voltage V OUT  to the load  490 . 
     The power switch Q 1  then turns off at the end of the primary interval D. The inductor current I LF  established during the primary interval D initially flows to the output, through the clamping capacitor C C , a body diode D 2  of the clamping switch Q 2  and the freewheeling diode D 1 . As the inductor current I LF  flows through the clamping capacitor C C , the clamping capacitor C C  begins to charge and a clamping voltage is developed thereacross. Since the clamping capacitor C C  and clamping switch Q 2  are coupled across the clamping inductor L C , the clamping voltage is applied across the clamping inductor L C , causing a clamping inductor current I LC  (which is a portion of the inductor current I LF ) to begin to flow through the clamping inductor L C . Then, as the clamping capacitor C C  continues to charge, a portion of the inductor current I LF  flowing through the body diode D 2  and the clamping capacitor C C  decreases while the clamping inductor current I LC  increases. Once the portion of the inductor current I LF  through the clamping capacitor C C  has decreased to zero, a substantial portion of the inductor current I LF  will flow through the clamping inductor L C  until the clamping switch Q 2  is turned on. 
     The clamping switch Q 2  is turned on for a short interval at the end of a complementary interval 1−D to apply the clamping voltage across the clamping inductor L C , causing the clamping inductor current I LC  to ramp up above the inductor current I F . The clamping switch Q 2  is then turned off. A difference between the inductor current I LF  and the clamping inductor current I LC  causes the voltage at the node  420  to rise rapidly to substantially the input voltage V IN . The power switch Q 1  may now be turned on with substantially zero volts thereacross to initiate the primary interval D. Conventionally, the turn-on of the power switch Q 1  would cause the freewheeling diode D 1  to exhibit reverse recovery currents for a short time. The present invention, however, advantageously positions the clamping inductor L C  in series with the freewheeling diode D 1  such that the clamping inductor current I LC  through the clamping inductor L C  effectively resists the current from the filter capacitor C F  thus reducing energy losses due to reverse recovery. 
     In the illustrated embodiment, the filter capacitor C F  is coupled across the output of the buck-boost converter  400 . With the filter capacitor C F  coupled as shown, the buck-boost converter  400  is operated with a discontinuous input current. While the discontinuous input current may be acceptable for a number of applications, some applications may be better served by a buck-boost converter operable with a continuous input current. 
     Turning now to FIG. 5, illustrated is a schematic diagram of another embodiment of a non-isolated buck-boost converter  500  constructed in accordance with the principles of the present invention. The buck-boost converter  500  is similar to the buck-boost converter  400  of FIG. 4 but includes a filter capacitor C F , coupled between an input and an output of the buck-boost converter  500 , to allow the buck-boost converter  500  to be operated with a continuous input current. 
     Turning now to FIG. 6, illustrated is a schematic diagram of an embodiment of a non-isolated Sepic converter  600  constructed in accordance with the principles of the present invention. The Sepic converter  600  has an input couplable to a source of electrical power  605  having an input voltage V IN  and an output couplable to a load  690 . The Sepic converter  600  includes a first filter inductor L F1  coupled to the input. The Sepic converter  600  further includes a power switch Q 1  coupled to the first filter inductor L F1 . The Sepic converter  600  further includes a capacitor C coupled to a first node  610  between the power switch Q 1  and the first filter inductor L F1 . The Sepic converter  600  further includes a second filter inductor L F2  coupled between the capacitor C and common. The Sepic converter  600  further includes a freewheeling diode D 1  coupled to a second node  620  between the capacitor C and the second filter inductor L F2 . The Sepic converter  600  further includes a filter capacitor C F  coupled across the output. 
     The Sepic converter  600  further includes an active clamp  630  coupled between the freewheeling diode D 1  and the second node  620 . In the illustrated embodiment, the active clamp  630  includes a clamping inductor L C  coupled in series with the freewheeling diode D 1 . The active clamp  630  further includes a series-coupled clamping capacitor C C  and clamping switch Q 2  coupled in parallel with the clamping inductor L C . The Sepic converter  600  still further includes a controller coupled to the power switch Q 1  and the clamping switch Q 2 . In the illustrated embodiment, the controller  650  monitors the output voltage V OUT  and controls the power and clamping switches Q 1 , Q 2  to regulate the output voltage V OUT  of the Sepic converter  600 . 
     The operation of the Sepic converter  600  is analogous to the operation of the buck converter  100  of FIG.  1  and the boost converter  300  of FIG. 3 and, as a result, will not be described. 
     Turning now to FIG. 7, illustrated is a schematic diagram of an embodiment of a non-isolated capacitive-coupled buck-boost converter  700  constructed in accordance with the principles of the present invention. The converter  700  has an input couplable to a source of electrical power  705  having an input voltage V IN  and an output couplable to a load  790 . The converter  700  includes a first filter inductor L F1  coupled to the input. The converter  700  further includes a power switch Q 1  coupled to the first filter inductor L F1 . The converter  700  further includes a capacitor C coupled to a first node  710  between the power switch Q 1  and the first filter inductor L F1 . The converter  700  further includes freewheeling diode D 1  coupled between the capacitor C and common. The converter  700  further includes a second filter inductor L F2  coupled to the capacitor C. The converter  700  further includes a filter capacitor C F  coupled across the output. 
     The converter  700  further includes an active clamp  730  coupled between the freewheeling diode D 1  and a second node  720  (between the capacitor C and the second filter inductor L F2 ). In the illustrated embodiment, the active clamp  730  includes a clamping inductor L C  coupled in series with the freewheeling diode D 1 . The active clamp  730  further includes a series-coupled clamping capacitor C C  and clamping switch Q 2  coupled in parallel with the clamping inductor L C . The converter  700  still further includes a controller  750  coupled to the power switch Q 1  and the clamping switch Q 2 . In the illustrated embodiment, the controller  750  monitors the output voltage V OUT  and controls the power and clamping switches Q 1 , Q 2 to regulate the output voltage V OUT  of the converter  700 . 
     The operation of the converter  700  is analogous to the operation of the buck converter  100  of FIG.  1  and the boost converter  300  of FIG. 3 and, as a result, will not be described. 
     Turning now to FIG. 8, illustrated is a schematic diagram of an embodiment of a non-isolated Zeta converter  800  constructed in accordance with the principles of the present invention. The Zeta converter  800  has an input couplable to a source of electrical power  805  having an input voltage V IN  and an output couplable to a load  890 . The Zeta converter  800  includes a power switch Q 1  coupled to the input. The Zeta converter  800  further includes a first filter inductor L F1  coupled between the power switch Q 1  and common. The Zeta converter  800  further includes a capacitor C coupled to a first node  810  between the power switch Q 1  and the first filter inductor L F1 . The Zeta converter  800  further includes a freewheeling diode D 1  coupled to the capacitor C. The Zeta converter  800  further includes a second filter inductor L F2  coupled to the capacitor C. The Zeta converter  800  further includes a filter capacitor C F  coupled across the output. 
     The Zeta converter  800  further includes an active clamp  830  coupled between the freewheeling diode D 1  and a node  820  (between the capacitor C and the second filter inductor L F2 ). In the illustrated embodiment, the active clamp  830  includes a clamping inductor L C  coupled in series with the freewheeling diode D 1  . The active clamp  830  further includes a series-coupled clamping capacitor C C  and clamping switch Q 2  coupled in parallel with the clamping inductor L C . 
     The Zeta converter  800  still further includes a controller  850  coupled to the power switch Q 1  and the clamping switch Q 2 . In the illustrated embodiment, the controller  850  monitors the output voltage V OUT  and controls the power and clamping switches Q 1 , Q 2  to regulate the output voltage V OUT  of the Zeta converter  800 . 
     The operation of the Zeta converter  800  is analogous to the operation of the buck converter  100  of FIG.  1  and the boost converter  300  of FIG. 3 and, as a result, will not be described. 
     Those skilled in the art should understand that the previously described embodiments of the active clamp and power supply are submitted for illustrative purposes only and other embodiments of the active clamp capable of mitigating the adverse effects of reverse recovery currents in the freewheeling diode and enabling substantially zero voltage switching of the power and clamping switches are well within the broad scope of the present invention. Additionally, exemplary embodiments of the present invention have been illustrated with reference to specific electronic components. Those skilled in the art are aware, however, that components may be substituted (not necessarily with components of the same type) to create desired conditions or accomplish desired results. For instance, multiple components may be substituted for a single component and vice-versa. 
     For a better understanding of power converters, see  Modern DC - to - DC Switchmode Power Converter Circuits,  by Rudolph P. Severns and Gordon Bloom, Van Nostrand Reinhold Company, New York, N.Y. (1985); and  Principles of Power Electronics,  by John G. Kassakian, Martin F. Schlect and George C. Verghese, Addison-Wesley Publishing Company, Reading, Mass. (1991). The above-listed references are incorporated herein by reference in their entirety. 
     Although the present invention has been described in detail, those skilled in the art should understand that they can make various changes, substitutions and alterations herein without departing from the spirit and scope of the invention in its broadest form.