Abstract:
An offset correction analogic circuit capable of compensating the offset of a digital baseband is described. The circuit comprises analog means to receive an output differential signal issued from a differential signal path. The differential signal path may be a baseband filter as used in digital communication systems. The baseband filter inputs an input differential signal having an offset to be compensated. The output differential signal is fed into an analog integrator to generate a pulsed signal either on a first output line or on a second output line according to the polarity of the output differential signal. The pulsed signal is then integrated into a switched capacitor and a differential compensation offset signal is issued. The frequency of the pulsed signal is preferably proportional to the voltage value of the output differential signal. The compensation offset signal may be summed with the input differential signal into a summing circuit.

Description:
TECHNICAL FIELD  
         [0001]    The present invention relates to offset compensation circuits, and more particularly to such circuits for baseband filters.  
         BACKGROUND ART  
         [0002]    Cellular telephones, as with most communication systems, require high gain baseband filters within the receive signal path. In such applications, the in-band signal is amplified and conveyed to subsequent stages for processing, e.g., to an analog-to-digital converter (ADC). This analog filtering serves two purposes: reducing the magnitude of interfering signals outside the band of interest; and providing anti-aliasing.  
           [0003]    The DC offsets in the receive signal path cause performance of the system to degrade in at least two ways. Offsets near the front end of the system get amplified by the active filter circuit and thereby reduce the available dynamic range of the ADC at the output. Additionally, offsets create errors in the two receiver signal paths commonly referred to as “in-phase” (I) and “quadrature” (Q) signal paths, thereby creating constellation distortion. Offset within the quadrature signal paths has been removed in conventional systems by using a low frequency feedback loop to cancel such offset component. In a conventional CDMA spread spectrum cellular telephone system, for example, the baseband information bandwidth extends from one to 1884 kilohertz (kHz). So as to not attenuate the low frequency baseband information, the offset cancellation loop bandwidth must be kept well below 1 kHz. So as to maintain signal integrity, low frequency phase response and group delay matching between the I and Q channels is just as important as magnitude matching. The offset cancellation loop bandwidth is typically set to approximately 100 hertz (Hz) to satisfy such requirements.  
           [0004]    The offset compensation generally may be solved by two conventional approaches. In the first one, the offset is detected by a digital baseband processor, and a feedback signal is provided by way of a pulse density modulated (PDM) output signal generated by a modulator/demodulator (MODEM) chip. This digital signal is filtered by a first order resistive-capacitive (RC) network. The output of this filter is next fed into an analog receive filter. The drawback of such solution is that it requires external components (the RC network) for filtering the output of the PDM. Moreover, some of the baseband digital processors do not include the offset compensation feature.  
           [0005]    The second prior art solution is based on a fully analogic feedback compensation system wherein the feedback signal is generated by way of analog integrator. As an example, U.S. Pat. No.5,471,665 from Pace and al. discloses a differential direct current (DC) offset compensation circuit for providing DC offset compensation to a circuit device. The DC offset compensation circuit comprises a differential integrator and a summing network. The circuit device is one of several types of devices, such as a DC coupled amplifier. The differential input of the circuit device is suitable for coupling to a differential input source and the differential output of the circuit device is suitable for connection to a load. The differential integrator features a transconductance amplifier at least one other amplifier, and a capacitor element having a capacitance of C1. The summing network sums the differential integrator output with the differential input signals of the differential input source and cancels DC offsets of the differential input source and the circuit device. Such solution either implemented using a transconductance amplifier associated to capacitors or implemented using a charge pump results in excessive area and not being integrable on-chip due to the high capacitance (of the order of several nanofarad) required to obtain the low bandwidth needed.  
           [0006]    Therefore there is a need for a fully analogic integrable offset compensation system.  
         SUMMARY OF THE INVENTION  
         [0007]    Accordingly, it is an object of the present invention to provide an offset correction analogic circuit capable of compensating the offset of a digital baseband.  
           [0008]    This object is achieved by employing a circuit which comprises analog means to receive an output differential signal issued from a differential signal path. The differential signal path may be a baseband filter as used in digital communication systems. The baseband filter inputs an input differential signal having an offset to be compensated. The output differential signal is fed into an analog integrator to generate a pulsed signal either on a first output line or on a second output line according to the polarity of the output differential signal. The pulsed signal is then integrated into a switched capacitor and a differential compensation offset signal is issued. The frequency of the pulsed signal is preferably proportional to the voltage value of the output differential signal. The compensation offset signal may be summed with the input differential signal into a summing circuit. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0009]    [0009]FIG. 1 shows a general block diagram of the present invention.  
         [0010]    [0010]FIG. 2 is a preferred implementation of a frequency modulated pulse generator used in the circuit of FIG. 1.  
         [0011]    [0011]FIG. 3 illustrates an hysteresis curve of the hysteresis comparators used in the circuit of FIG. 2.  
         [0012]    [0012]FIG. 4 is a preferred implementation of a switched capacitor used in the circuit of FIG. 1.  
     
    
     DETAILED DESCRIPTION OF THE INVENTION  
       [0013]    Referring now to the drawings, and more particularly to FIG. 1, a block diagram of the present invention is described. Generally speaking, the invention is preferably used in conjunction with a baseband signal path  100  that filters and gains an input differential signal ‘BB_in’ having an offset to be compensated. The output of the baseband filter  100  is a differential baseband output signal ‘BB_out’ to be used by an output load such as for example an A/D converter. A summing device  102  is connected in front of the baseband filter  100  allowing to sum an offset feedback output signal ‘OF_out’ generated by the system of the invention to the differential baseband input signal ‘BB_in’ in order to compensate the offset.  
         [0014]    The offset compensation system is a feedback loop connected between the output of the baseband filter  100  and the input of the summing device  102 . The feedback loop is made of a frequency modulated pulse generator  106  connected to a switched capacitor integrator  108 . The frequency modulated pulse generator senses the differential baseband output signal ‘BB_out’ to output a frequency modulated pulsed digital signal either on an ‘up-line’ or on a ‘down-line’ according to the polarity of the baseband output signal ‘BB_out’. The outputs of the pulse generator  106  are connected to respective inputs of the switched capacitor integrator  108  which output is connected to the summing device  102  either directly or through a low pass filter circuit  110 .  
         [0015]    The specific details of operation of the summing device  102  and of the gain and filter circuit  100 , which are well understood by those skilled in the art, will be omitted from this discussions. However, it is to be noted that if the baseband filter is an active RC filter the summing device may only consist of two resistors. Those alternatives have no impact on the invention.  
         [0016]    In operation, according to the polarity of the differential baseband output signal ‘BB_out’, the pulsed signal is generated by the pulse generator either on the up-line or on the down-line. The frequency of the pulses is proportional to the absolute value of the output baseband filter voltage ‘BB_out’. The pulses are next integrated by the integrator device  108  to generate the differential compensation offset signal ‘OF_out’. Each pulse generated on the up-line, respectively on the down-line, generates a positive step, respectively a negative step, at the output of the switched capacitor integrator  108 . Finally, the compensation offset signal ‘OF_out’ is summed to the baseband input signal ‘BB_in’ through circuit  102  in a well known manner to close the feedback loop, and the signal fed to the baseband filter is offset compensated.  
         [0017]    In an alternate embodiment, the output of the switched capacitor integrator  108  is fed into a low passed filter  110  to generate a smoothed ‘OF_out’ signal. The low pass filter may be implemented in the form of a common RC circuit.  
         [0018]    Referring now to FIG. 2, a preferred embodiment of the frequency modulated pulse generator  106  is detailed. The generator is composed of a continuous time integrator connected in series with a pair of differential hysteresis comparators  208 - a,    208 - b.  The time integrator is made of a pair of variable resistors  202 , each being connected in series on each differential baseband output line. Each second node of the series resistors is respectively connected to the input of a differential amplifier  206 . The differential output of this latter is input to a pair of differential hysteresis comparators  208 - a,    208 - b.  Those skilled in the art will readily understand that the input polarity of comparator  208 - a  is inverted to the input polarity of comparator  208 - b  to operate in opposite phase. The output of each comparator is the respective up-line and down-line generating the ‘up’ and ‘down’ signals to be inputted to the switched capacitor integrator  108 .  
         [0019]    A reset signal ‘Reset’ is generated by performing a logic OR function  210  between the up and down lines. This reset signal triggers two switches  204  respectively placed at the input and the output of the continuous time integrator to allow reset of this latter.  
         [0020]    Going now to FIG. 3, the principle of the hysteresis comparators  208 - a  and  208 - b  are now explained. FIG. 3 shows the hysteresis response of the comparators as used by the present invention. On the horizontal axis, the input voltage is represented while the output voltage is shown on the vertical axis. The two hysteresis high and low thresholds are referenced as Vth and Vtl, and is to be noted that a feature of the present invention is to fix both thresholds to a positive value in order to avoid that the circuit stays in a reset state indefinitely, as it will be the case for a negative value of the low threshold. The value of Vth depends on the saturation level of the continuous time integrator. The value of Vtl may be chosen in the range of 0 to Vth. However, the closer Vtl and Vth will be, the larger the bandwidth of the loop.  
         [0021]    It is to be appreciated that a major advantage of the invention is to adapt easily the bandwidth of the feedback loop. In fact, those skilled in the art will easily understand that the mean period ‘T’ between two pulses is equal to:  
               T   =         (     Vth   -   Vtl     )        RC     BB_out       ,           (   1   )                               
 
         [0022]    wherein R is the value of the variable resistor and C the capacitance of the differential amplifier  206 .  
         [0023]    From equation (1) it appears that the frequency of the pulses is proportional to the ‘BB_out’ signal voltage, and could be modified by simply adjusting the value of the two variable resistors  202 .  
         [0024]    It is to be noted that the above described preferred embodiment is not to limit the invention and that any other implementation could be adapted, such as for example a generation of the up and down signals directly made by a digital baseband processor.  
         [0025]    Moreover, the present invention is not limited to a particular technology process, and may be implemented using for example the MOSFET technology.  
         [0026]    Going now to FIG. 4, a preferred embodiment of the switched capacitor integrator  108  is now described. A first capacitor  402 - a  having one input connected to ground can either sample a first reference voltage Ref+ through a first switch  404 - a  or a second reference voltage Ref− through a second switch  406 - a,  by a corresponding connection of a second input. The opposite voltage references Ref+ and Ref− can be generated by conventional means such as a D/A Converter, or by any device generating a controllable differential voltage.  
         [0027]    Respectively, a second capacitor  402 - b  having one input connected to ground can either sample the second reference voltage Ref− through a third switch  404 - b  or the first reference voltage Ref+ through a fourth switch  406 - b,  by a corresponding connection of a second input. First and third switches  404 - a  and  404 - b  are controlled by the ‘down’ signal from the down-line, while second and fourth switches  406 - a  and  406 - b  are controlled by the ‘up’ signal from the up-line. The two capacitors  402 - a  and  402 - b  are furthermore respectively connected to a pair of fifth and sixth switches  408 - a  and  408 - b  to allow the transfer of the respective capacitors&#39; charges to integration capacitors  414 - a  and  414 - b  thanks to a differential amplifier  412 . The two switches  408 - a  and  408 - b  are triggered by a logic NOR device  410  obtained between the NOR function of the up and down signals. In operation, those skilled in the art will understand that when a pulse is sent on the up line, the differential voltage at the output of the integrator is modified according to equation (2):  
                 (       Ref   +     -     Ref   -       )     ×       C   i       C   s         ,           (   2   )                               
 
         [0028]    where Ci is the value of the integration capacitance ( 414 - a,    414 - b ) and Cs is value of the sampling capacitance ( 402 - a,    402 - b ). Respectively when a pulse is sent on the down line, the differential voltage at the output of the integrator is modified according to equation (3):  
               -     (       Ref   +     -     Ref   -       )       ×         C   i       C   s       .             (   3   )                               
 
         [0029]    It is to be recall that the output of the differential amplifier  412  is the differential compensation offset signal signal ‘OF_out’ of the offset compensation loop.  
         [0030]    In an alternate embodiment, the ‘OF_out’ signal may be freezed after a preliminary offset compensation operation by simply turning off the frequency modulated pulse generator.  
         [0031]    The skilled person will appreciate that the present configuration allows to easily modify the voltage variation at the output of the integrator  106  associated with each pulse, simply by controlling the voltage difference between the positive reference Ref+ and the negative reference Ref−.  
         [0032]    It is also to be noted that the noise at the output of the integrator  106  can be made compatible with the noise level required by the commonly used today standards such as the CDMA or the GSM ones at the input of their analog baseband.  
         [0033]    Finally, the advantages of the present invention are mainly that the switching frequency is proportional to the output signal thereby having the characteristics of a pseudo-random signal, contrary to common switched capacitors which generate undesirable noise.  
         [0034]    Moreover, it has been experimented in W-CDMA application that the total capacitance used by this solution has led to a total capacitor value of 20 picofarad.  
         [0035]    It will be apparent to those skilled in the art having regard to this disclosure that other modifications of this invention beyond those embodiments specifically described here may be made without departing from the spirit of the invention. Accordingly, such modifications are considered within the scope of the invention as limited solely by the appended claims.