Abstract:
A system for measuring damping and that includes an oscillator that produces an excitation signal for a resonator that can be placed in a damping medium. A sensor produces a sensor signal responsive to resonator motion. Also, a timing circuit ensures that excitation and sensing occur during mutually exclusive periods. An amplifier responds to the sensor signal, producing an amplified sensor signal. A phase detector is adapted to measure the phase relationship between the excitation signal and the amplified sensor signal and a controller is responsive to the phase detector to adjust the excitation frequency of the excitation signal, to create a phase lock loop. An integrator receives the amplified signal during periods that are mutually exclusive to and interleaved with the excitation. This integrator produces an integrated DC and low frequency component of the amplified signal, which is subtracted from the input amplifier input.

Description:
BACKGROUND 
     The present invention has to do with a method of measuring induced vibrations, over a sequence of periods, in a damped vibratory system and more particularly to the correction of an erroneous DC signal produced by an amplifier, together with low frequency interference, in such a system. One important application of this technology is the measurement of damping. 
     The measurement of damping, in a damped vibratory or resonant system, finds many application in the industrial arts, among the most important being the measurement of viscosity. One method of measuring viscosity, disclosed in U.S. Pat. No. 5,837,885, involves the perturbation of a fluid by a transducer vibrating near its resonant frequency. The excitation of the transducer is periodically stopped, and after a pause the transducer vibrations are measured. The phase of the received signal is logically “anded” with a phase-shifted signal, which is produced by adding a phase shift δΦ to the excitation signal, producing a control signal which is zero when the two signals are 90° separate in phase. The control signal is used to adjust the excitation frequency in the next iteration. After a number of iterations, typically on the order of 500, with each iteration taking about 1 millisecond, the phase of the received signal should be 90° different from the phase-shifted signal and the frequency should not change from one iteration to the next. The frequency at this state (phase-lock state) is measured. In order that a measurement of damping, and therefore viscosity, can be derived from this frequency, the phase is shifted in sequential periods, each of about one second, by δΦ; −δΦ, and then δΦ again, with the three resulting frequency values being used to determine damping, as explained further in the Detailed Description. The value of δΦ is typically 22.5° or 45°. 
     The patent discussed above represents a significant advancement in the art of viscosity measurement, with the method and apparatus disclosed gaining widespread acceptance in the field. Nevertheless, possible implementations of this method are limited to a set of applications, which it would be desirable to broaden, although already quite broad. 
     One set of problems is caused by a DC offset introduced by the amplifier used to amplify the sensed transducer signal. Every amplifier introduces some DC offset, however slight, into its output signal. This DC offset varies with time and temperature. For many electrical devices there is a tradeoff between accuracy and an added cost for expensive components that introduce less DC offset into the system. It is desirable to have a design that permits the use of less expensive components and yet returns a highly accurate result. The DC offset is a potential source of amplifier saturation with attending system nonlinearity, whereas minimizing the potential error caused by the DC offset adds to system complexity. 
     Unfortunately, the switching between excitation periods and sensing periods in the method of the &#39;885 patent makes it counterproductive to introduce a simple high pass filter into the system to filter out the erroneous DC voltage introduced by the amplifier. Transients that result from the switching process typically have frequency components of frequencies comparable to that of the vibration mode being measured, and thus are passed or even amplified by a conventional high-pass filter. Further complicating the task of reducing the DC offset is the fact that it is the DC offset specifically during the sense periods that should be corrected. Any method not timed to avoid being affected by the excitation period receive signal would risk corrupting the sense period measurement. Not only would such a method not effectively address the problem, but it could even make it worse. 
     One problematic condition is the measurement of fluid viscosity in an environment that includes a low frequency vibration. This condition occurs in many environments in which it is desired to measure viscosity, for example an industrial or treatment plant in which fluid is being pumped. The pump typically will introduce low frequency vibrations, which may interfere with the frequency measurement, leading to a less certain reading. There is even a possibility that such low frequency vibrations could defeat the phase-lock process, making it impossible to obtain a reading. Low frequency vibrations caused by a physical shock to the viscometer can have the same effect. 
     Accordingly, it would be desirable to have a method in which an erroneous DC signal produced by a system amplifier and low frequency vibrations caused by ambient noise or a sudden shock to the system could be reduced in amplitude. The frequent switching between excitation and sensing modes greatly complicates the task of originating such a system. 
     SUMMARY 
     In a first separate aspect, the present invention may take the form of a system for measuring damping and that includes an oscillator that produces an excitation signal for a resonator that can be placed in a damping medium. A sensor produces a sensor signal responsive to resonator motion. Also, a timing circuit ensures that excitation and sensing occur during mutually exclusive periods. An amplifier responds to the sensor signal, producing an amplified sensor signal. A phase detector is adapted to measure the phase relationship between the excitation signal and the amplified sensor signal and a controller is responsive to the phase detector to adjust the excitation frequency of the excitation signal, to create a phase lock loop. An integrator receives the amplified signal during periods that are mutually exclusive to and interleaved with the excitation. This integrator produces an integrated DC and low frequency component of the amplified signal, which is subtracted from the input amplifier input. 
     In a second separate aspect, the present invention may take the form of a method for measuring the damping of a damped vibratory system. The method includes periodically exciting the damped system with an excitation signal and alternately sensing the damped system response to the excitation signal and producing a sense signal at a sense signal node. Also, the sense signal is amplified to produce an amplified sense signal and the phase relationship between the excitation signal and the amplified sense signal is measured and the frequency of the excitation signal is adjusted so as to create a phase lock loop. Finally, a DC and low frequency component of the amplified sense signal is integrated, when the damped system is not being excited, to produce an integrated DC and low frequency signal, which is subtracted from the sense signal at the sense signal node. 
     In a third separate aspect, the present invention is a method of measuring properties of a fluid that uses a conductor electrically connected to a current source, so as to permit the current source to send a current through the conductor. A magnetic field is created about the conductor and the conductor is introduced into the fluid medium. A current waveform, having a frequency, is periodically passed through the conductor, so as to cause the conductor to move, due to force exerted on the conductor from interaction of the current and the magnetic field. The conductor movement is alternately sensed and a sense signal is produced at a sense signal node and is amplified to produce an amplified sense signal. The phase relationship between the current waveform and the amplified sense signal is measured and the current waveform frequency is adjusted so as to create a phase lock loop. The frequency when the phase lock loop is in lock state is measured as the phase between the excitation and the measured sense signal is varied, and fluid properties are calculated from the measured frequencies. 
     In a fourth separate aspect the present invention may take the form of an apparatus for measuring properties of a fluid or compliant solid, such as a gel. The apparatus includes a conductor electrically connected to a current source, so as to permit the current source to send a current through the conductor and a magnet placed to create a magnetic field about the conductor. A controller is connected to the current source and commands the current source to periodically send a current waveform, having a frequency, through the wire, so as to cause the conductor to move, due to force exerted on the wire from interaction of the current and the magnetic field. A sensor is adapted to sense the wire movement and produce a sense signal at a sense signal node and an amplifier is responsive to the sensor to amplify the sense signal to produce an amplified sense signal. Also, a phase detector is adapted to measure phase relationship between the current waveform and the amplified sense signal. The current source controller is responsive to the phase detector and is adapted to cause the current source to adjust the current waveform frequency so as to create a phase lock loop. Also, a frequency detector detects the current waveform frequency when the phase lock loop is in lock state and a logic unit calculates fluid properties from the measured frequency. 
     Other features of the present invention will be apparent from the accompanying drawings and from the detailed description that follows. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Exemplary embodiments are illustrated in the drawings. It is intended that the embodiments and figures disclosed herein are to be considered illustrative rather than restrictive. 
         FIG. 1  is a flow diagram of a method of determining damping in a damped vibratory system, which forms the environment of the present invention. 
         FIG. 2  is a functional block diagram of a preferred embodiment of the present invention in the form of a viscosity measuring device. 
         FIG. 3  is an illustration of the waveforms used in the device of  FIG. 2 . 
         FIG. 4  is a schematic diagram of the receive signal processing portion of the device of  FIG. 2 . 
         FIG. 5  is a graph of the amplified sensed signal response of a system according to the present invention to an oscillating input signal. 
         FIG. 6  is a graph of the prior art system that does not include the DC offset and low frequency integrator of the system of  FIG. 5 , with a DC offset added to cancel the DC signal produced by the system, but where this added DC offset is deliberately set too small to fully cancel this DC signal. 
         FIG. 7  is a graph of the response of the system of  FIG. 6 , with a DC offset added to the signal to cancel the DC offset produced by the signal amplifier, but where this added signal is deliberately slightly too large. 
         FIG. 8  is a graph showing 5 traces, each showing a response of the system of  FIG. 6  to a signal that includes 120 Hz noise at 100 mV. 
         FIG. 9  is a graph showing 5 traces, each showing a response of the system of  FIG. 6  to a signal that includes 120 Hz noise at 2 Volts. 
         FIG. 10  is a graph showing 5 traces, each showing a response of the system of  FIG. 5  to the same signal that produced the traces of  FIG. 9 . 
         FIG. 11  is a graph showing 5 traces, each showing a response of the system of  FIG. 5 , to the same signal that produced the traces of  FIG. 10 , except for that the DC and Low Frequency Integrator is activated a period earlier than in  FIG. 10 . 
         FIG. 12A  is a perspective cut-away view of a preferred embodiment of a transducer configuration. 
         FIG. 12B  is an exploded view of the transducer configuration of  FIG. 12A . 
         FIG. 12C  is a cross-sectional view of the transducer configuration of  FIG. 12A , taken along view line C-C. 
         FIG. 13  is a cross-sectional view of a transducer configuration from the prior art, but which could be used in the system of the present invention. 
         FIG. 14A  is an illustration of an additional alternative preferred embodiment of a transducer configuration. 
         FIG. 14B  is an additional illustration of the transducer configuration of  FIG. 14A , showing the direction of magnetic flux. 
         FIG. 14C  is another additional illustration of the transducer configuration of  FIG. 14A , showing torsion of the wire loop. 
         FIG. 14D  is still another additional illustration of the transducer configuration of  FIG. 14A , showing maximum torsion of the wire loop. 
         FIG. 15A  is an illustration of another additional alternative preferred embodiment of a transducer configuration. 
         FIG. 15B  is an additional illustration of the transducer configuration of  FIG. 15A , showing the direction of magnetic flux. 
         FIG. 15C  is another additional illustration of the transducer configuration of  FIG. 15A , showing planar distension of the wire loop. 
         FIG. 15D  is still another additional illustration of the transducer configuration of  FIG. 15A , showing maximum planar distension of the wire loop. 
         FIG. 16A  is an illustration of another additional alternative preferred embodiment of a transducer configuration. 
         FIG. 16B  is an additional illustration of the transducer configuration of  FIG. 16A , showing the direction of magnetic flux. 
         FIG. 16C  is another additional illustration of the transducer configuration of  FIG. 16A , showing bending of the plane of the wire loop. 
         FIG. 16D  is still another additional illustration of the transducer configuration of  FIG. 16A , showing maximum bending of the plane of the wire loop. 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     The present invention may be instituted in the environment of a damped-vibratory-system damping-measurement process  10  shown in flowchart form in  FIG. 1 . A transducer is used to periodically excite a resonator immersed in a fluid medium (block  12 ), by vibrating the resonator at an excitation frequency. Alternately, after a pause (block  14 ), the movement of the resonator is sensed by the transducer (block  16 ). After this, the phase of the sensed signal relative to the excitation signal phase, plus a phase shift 90°+δΦ, is computed (block  18 ). In one preferred embodiment this process is repeated for a fixed number of iterations, typically on the order of 500. In this embodiment, the next step is to determine if the fixed number of iterations has been reached (decision box  20 ). If the iteration count is not complete, the frequency is adjusted (block  22 ) in a manner designed to drive the phase difference to 90°+δΦ and the process is repeated (starting with block  12 ). 
     After repeating this process for the fixed number of sense period iterations the final frequency is recorded (block  24 ). Next, the process is repeated for −δΦ, and the frequency corresponding to −δΦ is recorded (block  24 ). Absent temperature shift during sensing, the phase lock frequency for phase shift δΦ minus the phase lock frequency for phase shift −δΦ, yields damping. To correct the final reading for possible temperature drift, however, the process is run again for +δΦ with the two resultant final frequencies for +δΦ being averaged. Finally, damping is computed from the difference between the averaged frequency value for +δΦ and the frequency value for −δΦ (block  28 ). In the case where fluid viscosity is being measured, viscosity is computed from the measured damping value. 
     Referring to  FIGS. 1 ,  2  and  3 , in one preferred embodiment the present invention takes the form of a viscometer system  110  in the form of a phase lock loop receiver. System  110  uses a sensor (two configurations of which are shown in  FIGS. 12A-12B  and  13 A- 13 B and described in the accompanying text), composed of a resonator and a transducer  114 . Transducer  114  is used to vibrate (excite, block  12 ) a resonator and, after a pause (block  14 ), sense (block  16 ) the residual vibration. The transducer  114  is driven by excitation signal  210  ( FIGS. 2 and 3 ), which is the output  211  ( FIG. 2 ) of a voltage controlled oscillator (VCO)  115  which is switched on and off by switch S 3  controlled by an excitation gate signal  212  ( FIG. 3 ), produced by an excitation gate generator  116 . 
     The sensed signal  214 , minus the output  240  of the DC and low frequency signal integrator  120  (described below) drives an amplifier  122 , to produce an amplified sensed signal  223  (location shown in  FIG. 2 ). In alternation with the excitation gate signal  212 , after a time gap to permit transients produced by the excitation to attenuate, a sense gate signal  216 , produced by a sense gate generator  118 , changes state causing the phase of the amplified sensed signal  223  to be processed in order to supply a frequency correction signal (block  18 ). This measurement is performed by an in-phase integrating synchronous detector  130 , which determines the phase shift of signal  223  relative to the phase of a phase-shifted signal  220 , produced by applying a phase shifter  124  to the VCO output  211 . Phase-shifted signal  220  is gated by the sense gate signal  216  to create a reference signal  221 . A quadrature integrating synchronous detector  132  performs the same task for the quadrature portion of the signal, which is used to drive the automatic gain control (AGC)  134 . 
     The in-phase detector  130  produces a frequency control signal  222  that is proportional to the integrated measurement of the phase difference of the amplified sensed signal  223  and the phase shifted reference signal  220 , which produces a zero result when signals  223  and  222  are 90° out of phase. This frequency control signal  222  in turn drives the VCO  115 , so that the excitation frequency produced by the VCO  115 , and applied during the next excitation gate will remain unchanged only when the phase difference between signals  223  and  220  is 90°, resulting in no change to the integrated phase difference value. The quadrature detector  132  drives the Automatic Gain Control mechanism  134 , which adjusts the signal strength of the voltage controlled oscillator, to maintain the signal strength in the dynamic range of the system  112 . 
     After it settles, the final frequency that results in the specified phase shift δΦ is recorded. As noted previously, in a preferred embodiment the process is permitted to run through a −δΦ iteration and an additional +δΦ iteration, the two frequencies for the +δΦ iterations are averaged together to correct for the effect of possible temperature variation, and the frequency for the −δΦ iteration subtracted from this average, to yield a quantity related to damping, from which a logic unit  142  ( FIG. 2 ) calculates damping (block  28 ). The measurement of damping, in turn, yields a measurement of fluid viscosity. 
     In the innovation of the present invention, a DC and low frequency signal integrator  120  is introduced into the circuit. Integrator  120  is driven by the output of the amplifier  122  during the sense gate signal  216 , when a switch S 1  is closed. When S 1  is open the input voltage to integrator  120  is held constant by a capacitor within unit  120 . The integrator  120  output is subtracted from the input of the amplifier  122 , by adder  121 . Because low frequency signals change slowly relative to the response time of the amplifier  122 , all low frequency signals from the transducer are driven to zero at the input (and therefore the output) of amplifier  122 . Any DC signal produced by amplifier  122 , itself, results in a DC signal of opposite polarity and correct amplitude to negate the DC signal that would otherwise appear at the output of amplifier  122 . 
     Referring to  FIG. 4 , which shows one embodiment of the in-phase signal processing portion of the viscometer  110  in greater schematic detail, integrator  120  is made up of an operational amplifier OA 1 , an integrating capacitor C 1  and resistor R 6 . The amplifier  122  is made up of a first operational amplifier OA 2  driving a second operational amplifier OA 3 , with resistors R 1 , R 2 , R 4  and R 5  determining the amplification. Resistors R 1  and R 7  serve as adder  121 . In an alternative topology operational amplifier OA 2  is included in adder  121 . 
     In-phase detector  130 , includes an analog switch S 2  that passes the amplified signal  223  when the reference signal  221  is high, to create a demodulated signal  224 . An operational amplifier OA 4 , resister R 8  capacitor C 3  form an integrator  140  for this signal, and this drives the VCO  115 . When reference signal  221  is exactly 90° out of phase with amplified sensed signal  223 , demodulated signal  224  integrates to zero. This is because under this condition signal  221  transitions from zero to one at the peak of signal  223  and then transitions from one to zero at the negative peak of signal  223 , so that a perfect half-cycle of signal  223 , with the negative quarter-cycle canceling the positive quarter-cycle, is sent to the integrator  140 . When this relationship occurs, the output of the demod integrator  140  will not change from one iteration to the next, and the system  110  will be in phase-lock state. 
     Returning to the integrator  120 , skilled persons will recognize that the time constant, and therefore the frequency response, of integrator  120  is set by the values of the capacitor and resistor, C 1  and R 6 , respectively. The value of 1/(2π*R 6 *C 1 ) sets the 3 db point of the circuit, at which an input signal is attenuated by a factor of 0.5 from its input value. Accordingly/if the product C 1 *R 6  of these two values were set equal to 0.02, for example, the 3 db frequency would be approximately 8 Hz. This is a realistic value for this type of circuit because many physical vibrations representing ambient noise would be in the 1 Hz range. A value of 100K for the resistor and of 0.2 μF for the capacitor would yield this value, and represent reasonable values for the circuit. Clearly, however, the value can be set over a range and depending on the application, it may be desirable to set the 3 dB attenuation point to anywhere from 5 Hz to 10,000 Hz. In one preferred embodiment the 3 dB point is set at about 100 Hz to reject ambient noise caused by rotating machinery. Other values outside of this range may be warranted for some applications. 
       FIGS. 5-11  are graphs that collectively show system performance improvements obtained by the introduction of integrator  120 .  FIGS. 5-7  taken together illustrate the effect of partially uncompensated DC offset.  FIG. 5  shows an oscilloscope trace of the received amplified signal  223  during normal operation of system  110 . For experimental purposes, however, a prior art system identical to that of system  110 , except for the absence of integrator  120  is used and a DC signal is introduced into the inverting input of operational amplifier OA 2  (using the same reference numbers for the prior art system). This signal is connected to the inverting input of OA 2  through a 52 kΩ resistor and is calibrated by being hand adjusted until the amplified signal  223  appears as it does in  FIG. 5 , yielding an offset compensation value of 2.497 volts. To determine the effect of not fully compensating for the DC offset, the introduced DC signal is adjusted so that it is 14 mvolts lower than the 2.497 volt compensation value, yielding the graph of  FIG. 6 , in which amplified signal  223  is pulled negative during the sense gate. Also, because of the partially uncompensated DC offset the automatic gain control (AGC) is misdirected, causing an increase in signal amplitude.  FIG. 7  shows the response of signal  223  to an injected voltage that is higher than the compensation voltage by 3 mvolts, driving the signal up and misdirecting the AGC in the opposite manner to that of the  FIG. 6  case, thereby reducing the signal amplitude. When the conditions of  FIGS. 6 and 7  are repeated for system  110 , the resultant amplified signal looks essentially identical to that of  FIG. 5 , as the integrator  120  completely compensates for the injected signal. In fact, even when the injected voltage differs from the offset compensation value by 2 volts, there is no discernable effect on the trace of signal  223 . 
       FIGS. 8-11  illustrate the effect of different integrator  120  time constants in filtering out 120 Hz input noise of varying amplitudes.  FIG. 8 , shows the amplified signal  223  of system  112 , when the time constant τ for the integrator  114  is 5×10 −2  and 100 mvolts, 120 Hz noise has been injected through a 1 MΩ resistor to the non-inverting input of operational amplifier OA 2  ( FIG. 4 ). In  FIG. 8  and the following graphs the input frequency equals the resonator third harmonic (@19,444 Hz), which will be discussed in greater detail below. Five superimposed traces are shown. Although the added noise is evident in the greater variation of the traces, no significant detriment is caused.  FIG. 9 , shows the response of a system identical to that of  FIG. 8 , but where the 120 Hz input noise has an amplitude of 2 volts, simulating noise that might be introduced into a fluid being measured by, for example, a pump. Five separate traces are shown superimposed. In this case the time constant of integrator  120  is not low enough to prevent the system response from being severely degraded by the noise, so that although phase lock is maintained, signal  223  is not at all consistent from one trace to the next, causing large errors in the measured damping. When the amplitude of the input noise is raised to 3 volts, phase lock is lost and the system ceases to function.  FIG. 10  shows the system response to the exact same signal input conditions, but this time with the integrator time constant decreased to 5×10 −4 . By reducing the value of capacitance by a factor of 100 the range of frequency addressed by the integrator is increased by the same factor. As can be seen, the response during the time the integrator  120  is active quickly becomes uniform between iterations.  FIG. 11  shows the same system and waveform input as  FIG. 10 , but in this case the integrator is switched on a period before the sense gate begins, during the dead time between excitation and sensing. By the time the sense gate switches on, the waveform is completely stabilized, eliminating the error due to the simulated vibrational disturbance. 
       FIGS. 12A-12C  show a cross-sectional view of viscometer transducer  114 . A coil  302  is used to apply magnetic force to a resonator  303 , made up of a resonator permanent magnet ( 304 ), inner tube  306 , and outer tube  308 , which is attached to inner tube  306  at the end furthest from magnet  304 . The resonator permanent magnet  304  is driven by the electromagnet flux created by coil  302  and in turn physically drives inner tube  306 , which physically drives outer tube  308 . The direction of the current through coil  302  alternates at the fundamental frequency, causing the resonator  303  to move rotationally back and forth at the fundamental frequency. The viscosity of the fluid being measured damps the motion of the outer tube  308 . It is this damping that is measured by system  110 , in order to form a measurement of viscosity. The coil  302  is held in place within a sensor body  312  and vibrationally isolated from body  312  by O-rings  310 . Sensor body  312  includes threads  314  to facilitate the placement of transducer  114  in the side of a pipe or tank. 
     Referring to  FIG. 13 , in a prior art transducer configuration  114 ′, sensor body  312 ′ is held in place in an outer housing  320  by a pair of rubber sensor-mounting O-rings  316 , one of which is held in place by a sensor retaining ring  318 . O-rings  316  are necessitated by the need to run prior art viscometers at the resonator fundamental frequency. As explained further below, the improvements to signal processing yielded by the system  110  make practical transducer  114 , which does not include sensor mounting O-rings  316 . The elimination of the rubber O-rings  316  makes transducer  300  more compact, easier to clean, less vulnerable to chemical and thermal attack and more robustly sealed. Accordingly, the newly practical transducer  300  expands the applications of viscometers. 
     The O-rings  316  act to isolate outer housing  320  from the vibrations of outer resonator tube  308 , which could otherwise cause parasitic resonances in a structure to which transducer  114 ′ was attached, when resonator  303  is run at its fundamental frequency. A system in which parasitic resonances may distort the measurement is unreliable and therefore impractical. But the possibility that low frequency noise would cause the phase-lock loop system to break phase lock prevented the use of higher resonator vibrational modes in prior art systems. 
     Because the DC and low frequency integrator  120  prevents low frequency noise from causing the system to break lock on a higher frequency, driving the resonator at its second torsional mode (whose frequency is typically 3 times that of its fundamental mode), becomes practical in system  110 . The second torsional mode, however, causes the outer tube vibration to apply less force to transducer body  312 , by dividing the vibration into three zones, each of which has a lower amplitude and applies less force to its zone endpoints than vibration in a single zone, at the fundamental frequency. Accordingly, the possibility of parasitic resonances is greatly reduced. 
     Referring to  FIG. 14A , in an additional preferred embodiment the function of transducer  114  is performed by specific transducer configuration  408 , which includes an elastic conductive hairpin loop  410 . A measurement of the damping of induced vibrations in this loop  410 , as performed by system  110 , can be used to determine the properties of a fluid in which the loop has been introduced. These properties are not limited to fluid viscosity, but include density and elasticity. In addition, a system  110 , equipped with transducer  408  can be used to measure the damping and elasticity of gels and other mechanically compliant solids 
     In one preferred embodiment, the loop  410  is made of an elastic wire, made of a metal, such as copper, stainless steel or silver. In alternative preferred embodiment, however, similarly loop-shaped conductors are composed of other materials fabricated in forms other than wire, for example, from etched, selectively conductive silicon, or from insulators such as ceramics or glass made conductive by metallization or other processes. 
     The loop  410 , all of which is conductive and elastic, includes a first leg  412 , a second leg  414  and a bridge  416  joining the two. Also, a massive, electrically insulating base  418  supports loop  410 . A current source  420 , drives a current through the loop  410 . Additionally, a pair of magnets  422  create a magnetic field that is traversed by loop  410 . Accordingly as current is passed through loop  410 , it is pushed by a mechanical force proportional at each point to the vector product of the magnetic field (the direction of which is shown by the letter “B” and associated arrow, a circle with a dot in the middle indicating an arrow pointing out of the sheet and a circle with an “x” inside indicating an arrow pointing into the page) and the current (represented by the letter “I” and associated arrow) through that segment, causing a mechanical distortion (represented by the letter “F” and associated arrow) of loop  410 . As the current direction in legs  412  and  414  is mutually opposed, this creates opposite forces in legs  412  and  414 , acting to twist loop  410 , as shown in  FIGS. 14B ,  14 C and  14 D. Furthermore, because the material of the loop  410  possesses both inertia due to its mass, and elasticity, when the loop  410  is distorted and released, it will vibrate at one of its characteristic frequencies, thereby having a set of vibratory modes. If the current source produces an alternating current, its frequency may be adjusted such as to preferentially excite one of the resonant modes of the wire loop. The conventions used in  FIGS. 14A-14D , to show magnetic field direction (B), current direction (I) and resultant mechanical force (F), are also used in  FIGS. 15A-15D  and  FIG. 16A-16B . 
     Referring to  FIG. 15A , a transducer configuration  408 ′ includes magnets  422 ′ that are positioned behind and in front (not shown) of the loop  410 , with opposing poles facing each other. The resulting magnetic field is perpendicular to the plane of the loop  410 , so that the force on the legs  412  and  414  of the loop is in the plane of the loop  410 , and either inward toward its symmetry axis, or outward away from its symmetry axis depending on the polarity of the current source  420 . The legs of the loop function like the tines of a tuning fork in which the tips of the tines are connected by an elastic member. This is shown in  FIGS. 15B ,  15 C and  15 D.  FIG. 15B  illustrates the magnetic, current and force vectors operating on the loop  410 .  FIG. 15C  shows the static distortion of the loop  410  compared to its initial undistorted shape.  FIG. 15D  shows the limits of the motion of the loop  410  when it is driven by magnets  422 ′ as shown in  FIG. 15A . 
     Referring  FIG. 16A , a transducer configuration  408 ″ includes a magnet  422 ″ that is positioned above the arch  416  of the loop  410 . Referring to  FIG. 16B , the magnetic field of magnet  422  is parallel to the plane of the loop and parallel to the current flow in its legs  412  and  414 , with only bridge  416 , which is not parallel to the field, experiencing a force, resulting in bridge  416  being pushed into and out of the paper, as shown in  FIG. 16C  and with  FIG. 16D  showing the limits of the motion of the loop when driven in the transducer configuration  408 ″. 
     When the loop is immersed in a fluid or compliant solid, each of the transducer configurations  408 ,  408 ′ and  408 ″ produces a somewhat different pattern of flow or distortion in the medium. The vibrational characteristics created by the differing transducer configurations  408 ,  408 ′ and  408 ″ will be influenced to differing degrees by the characteristics of the medium. Therefore, the transducer configuration, and therefore the vibratory frequencies, can be selected to separate the effects of various properties of the medium. Moreover, in additional preferred embodiments the magnets are not oriented the along a principal axis of the loop, as is shown in  FIGS. 14A ,  15 A and  16 A. In one preferred embodiment of a transducer configuration, the magnetic field is oriented at a nonzero angle to each of the principle axes, permitting all of the mode geometries cited above to be generated by a single transducer configuration. 
     Skilled persons will recognize that many other permutations of wire-based fluid parameter measurement devices are possible. For example, a straight wire could be used and would be subject to force when current flowed through it, similar to a leg of one of the above-described embodiments. Another example of the many possible geometries, would be a wire that includes a sharp bend at its distal end, which could be used to broach a membrane covering a fluid. Also, lithographic manufacturing techniques could be used to create a loop having a precisely defined geometry. 
     Many variant embodiments also exist for the arrangement of the magnets. For example, although two magnets are shown in the embodiments of  FIGS. 14A and 15A , a single magnet could suffice to create the required magnetic field, for the wire geometries shown. Alternatively, a magnetic loop could be used to create the required magnetic field. 
     While a number of exemplary aspects and embodiments have been discussed above, those of skill in the art will recognize certain modifications, permutations, additions and sub-combinations thereof. It is therefore intended that the following appended claims and claims hereafter introduced are interpreted to include all such modifications, permutations, additions and sub-combinations as are within their true spirit and scope.