Abstract:
A multi-tone signal amplifier topology and an amplifying method in which a first amplifier outputs a first signal having at least one fundamental frequency signal and a first distortion signal. A second amplifier outputs a second signal that has a fundamental frequency signal corresponding to each fundamental frequency signal of the first signal and a second distortion signal. Each fundamental frequency signal of the second signal is substantially in-phase with the corresponding fundamental frequency signal of the first signal, while the second distortion signal is substantially 180° out-of-phase with the first distortion signal. An output coupler combines the first and second signals to form a third signal having the corresponding fundamental frequency signals of the first and second signals constructively combined and a third distortion signal that is a difference between the first distortion signal and the second distortion signal. A detector, coupled to the third signal, generates a control signal applied to the second amplifier for adjusting at least one of an amplitude and a phase of the second distortion signal thereby minimizing the third distortion signal. The second amplifier circuit includes an adjust circuit that varies at least one of an amplitude and a phase of the input signal for the second amplifier in response to the control signal.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to radio frequency (RF) amplifiers. More particularly, the present invention relates to a multi-tone amplifier and a method for amplifying having adaptive closed-loop control for minimizing intermodulation distortion products, and for minimizing amplifier performance degradation caused by component drift, temperature variation and aging. 
     2. Description of the Related Art 
     As is well-known, when a dual or multi-tone input signal is applied to an amplifier that is not perfectly linear, undesirable intermodulation (IM) products are generated at predictable frequencies causing intermodulation distortion (IMD). Amplifiers operating in class AB or class B modes tend to produce high levels of IMD product when multi-frequency signals—that is, multi-tone signals—are amplified. IM product levels on the order of −30 dBc (30 decibels below the fundamental frequency or carrier level) are typical. The undesirable IM products are particularly apparent when the amplifier is operated in saturation or in the gain compression region of the amplifier. The level of the IM products are greater the further into the gain compression region the amplifier is operated. 
     Harmonic IM products are not of primary concern because they can be removed by a filter. Third order and fifth order IM products, however, fall within the desired communication bandwidth of the amplifier and cannot be removed by a filter. The only way to deal with the third and fifth order intermodulation products is to amplify in a way that does not generate third and fifth order intermodulation products. 
     A conventional technique for reducing intermodulation distortion (IMD) is to use a correction amplifier that generates correction signals at the same frequencies as the undesirable intermodulation (IM) products, but having phases that are 180° out-of-phase from the phases of the corresponding IM products. When the IM products and the correction signals are applied to an output combiner, the IM products are cancelled by vector summation with the correction signals. As a result, the amplified output signal has substantially only the fundamental input signal frequencies, i.e., the multi-tone components of the input signal. 
     FIG. 1 shows a schematic block diagram of a conventional low-distortion RF amplifier circuit  10  that includes a correction amplifier. Circuit  10  linearly amplifies an input signal S IN  to produce an amplified output signal S OUT . Input signal S IN  is a dual-tone high-frequency signal having sinusoidal components at a first fundamental frequency f1 and at a second fundamental frequency f2. For this description, frequency f2 is greater than frequency f1. Both frequencies f1 and f2 are within standard wireless communication frequency bands, such as between 800-960 MHz. The phase of S IN  is arbitrary. Amplification causes IM products to occur at both higher and lower frequencies than the communication frequency band of interest. 
     In FIG. 1, frequency components f1 and f2 of S IN  and various other signals are shown vectorally for conveniently showing phase relationships between the same frequency components at specific points within circuit  10 . Power and voltage standing wave ratio (VSWR) losses are ignored in the following description. 
     Input signal S IN  is applied to an input port  11  of a first coupler, or power splitter, C 1 . Coupler C 1  splits signal S IN  into a signal S 1  that is output at a “direct path” output port  12  and a signal S 2  that is output at a “coupled path” output port  13 . Typically, coupler C 1  is a passive device, such as conventional branch line coupler or Wilkinson-type divider, that splits input power unequally between output ports  12  and  13 , with higher power being output at port  12 . 
     Signal S 1  includes sinusoidal components at frequencies f1 and f2 having respective voltage levels of C 11 V 1  and C 11 V 2 , where C 11  is the coupling coefficient of coupler C 1 . The phases of the f1 and f2 components of signal S 1  are defined to be 0°. Similarly, signal S 2  includes sinusoidal components at frequencies f1 and f2 having respective voltage levels V 1 {square root over (1+L −C 11   2 +L )}, and having respective phases also defined to be 0°. 
     Signal S 1  is applied to a power amplifier A 1  where it is amplified to produce an amplified signal S 3  output at an amplifier output port  14 . Amplifier A 1  is a conventional high-frequency amplifier operating in class A, AB or B; for example, a power gain on the order of 30 dB to produce RF output power of about 50 W. 
     Amplified signal S 3  contains amplified frequency components f1 and f2 and undesirable intermodulation distortion products at frequencies f3 and f4. Frequency f3 is at 2f1−f2, and is less than frequency f1. Frequency f4 is at 2f2−f1, and is greater than frequency f2. The components of signal S 3  at frequencies f1 and f2 are G1C 11 V 1  and G1C 11 V 2 , respectively, where G1 is the voltage gain of amplifier A 1 . The phases of the f1 and f2 components of S 3  are −φ 10  and −φ 20 , respectively, where −φ 10  and −φ 20  are the respective insertion phase lags through amplifier A 1  at frequencies f1 and f2. The minus sign indicates a phase lag or delay. The intermodulation distortion components of signal S 3  at frequencies f3 and f4 have respective voltage levels V 3  and V 4  with respective reference phase values of −φ 30  and −φ 40 . 
     Signal S 3  is applied to an input port  15  of a coupler C 2 , such as a conventional hybrid (e.g., branch line), a backward firing or a Wilkinson coupler. Coupler C 2  has a coupling coefficient of C 22  that is typically in the range of −10 to −20 dB. A coupled-path signal S 4  is output from a coupling port  16  and is, for example, 10 to 20 dB below the level of a direct-path signal S 8  that is output from a direct port  17 . The voltage levels of the frequency components of signal S 4  are each C 22  times the corresponding voltage levels of the signal S 3  frequency components. The voltage levels of the components of signal S 8  are {square root over (1+L −C 22   2 +L )} times the corresponding voltage levels of the components of signal S 3 . The respective phases −φ 11  to −φ 41  of the frequency components f1-f4 of signal S 4  are the same as the phases of the corresponding frequency components of signal S 8 . Specifically, phase values −φ 11  and −φ 12  are the combination of the insertion phase lags −φ 10  and −φ 20  through amplifier A 1 , respectively, plus the respective insertion phase lags at frequencies f1 and f2 through coupler C 2 . Phase values −φ 31  and −φ 41  are the insertion phase lags at the respective frequency f3 and f4 through coupler C 2 , plus the phase lag through amplifier A 1 . 
     Coupled-path signal S 4  is applied to a phase shifter  18 , such as a variable capacitor-type phase shifter, PIN diode phase shifter or a Shiffman phase shifter, for introducing a 180° phase shift at each of the frequencies f1-f4. A signal S 5  output from phase shifter  18  is input to a coupled port  19  of a coupler C 3 . Signal S 5  contains the same frequency components f1-f4 at the same voltage levels as signal S 4 , but with the phase of each respective component shifted by 180° from the corresponding components of signal S 4 . Specifically, the voltage levels of the f1 and f2 components of signal S 5  are C 22 G1C 11 V 1  and C 22 G1C 11 V 2 , respectively, and the respective phases are −φ 11 −180° and −φ 21 −180°. The voltage levels of the f3 and f4 components of signal S 5  are C 22 V 3  and C 22 V 4 , respectively, and the respective phases are −φ 31 −180° and −φ 41 −180°. 
     Signal S 2  is input to a delay line DL 1 , which outputs a signal S 6 . Signal S 6  is input to a port  20  of coupler C 3 . Delay line DL 1  introduces phase lags of −φ 11  and −φ 21  at respective frequencies f1 and f2, equalling the insertion phase lags through amplifier A 1  plus coupler C 2  at frequencies f1 and f2. Thus, the f1 and f2 frequency components of signal S 6  are 180° out-of-phase with the f1 and f2 frequency components of signal S 5 . 
     Coupler C 3  substantially subtracts signal S 5  from signal S 6  to produce a signal S 7  having signal components f1-f4. In this case, the f1 and f2 components of signal S 7  have respective phase values that are equal to the phase of f1 and f2 components of signal S 5  or S 6 , depending on the cancellation in coupler C 3 . The voltage levels of the f1 and f2 components of signal S 7  are ({square root over (1+L −C 22   2 +L )} {square root over (1+L −C 22   2 +L )}−C 33 C 11 C 22 G1) times V 1  and V 2 , respectively, where C 33  is the coupling coefficient of coupler C 3 . Coupler C 3  also produces the f3 and f4 components of signal S 7  at voltage levels of C 33 C 22 V 3  and C 33 C 22 V 4 , respectively, with respective phase values of −φ 32  and −φ 42 . 
     Signal S 7  is applied to the input of an amplifier A 2 . The respective voltage or power levels of signal S 7  at frequencies f1 and f2 are below the corresponding power levels of signal S 1  as a function of the cancellation in coupler C 3 . Amplifier A 2  outputs a signal S 9 . The gain G2 of amplifier A 2  is selected such that the f3 and f4 terms cancel at the output. 
     Signal S 9  also contains distortion components at distortion frequencies f3 and f4, respectively designated as “f3,S 9 ” and “f4,S 9 ”, that are primarily the result of the amplification of the corresponding distortion frequency components of signal S 7 . A signal S 10 , which is signal S 8  delayed by a delay line DL 2 , contains distortion frequency components at respective frequencies f3 and f4 and respectively “f3,S 10 ” and “f4,S 10 ”. Signal S 10  also contains fundamental frequency components “f1,S 10 ” and “f2,S 10 ” at respective frequencies f1 and f2. 
     Output combiner  22  adds the f1 and f2 frequency components when the respective phases of f1,S 9  and f2,S 9  equal the corresponding phases of f1,S 10  and f2,S 10 , and when the corresponding voltage levels of S 9  and S 10  are equal. Delay line DL 2  equalizes the phases by adding an insertion phase lag that equals the insertion phase lag through amplifier A 2  and through the path between ports  20  and  21  of coupler C 3 . Delay line DL 2  also compensates for the insertion phases of the distortion products of signal S 5  through coupler C 3  and for the distortion products of signal S 7  through amplifier A 2  at f3 and f4. 
     Output combiner  22  receives signals S 9  and S 10  at ports  23  and  24 , respectively. Combiner  22 , such a 3 dB Wilkinson-type coupler, cancels the distortion frequency power within signals S 9  and S 10  adding the two signals 180° out-of-phase. Similarly, when two equal amplitude, but in-phase components are applied to ports  23  and  24 , all of the power appears at the output port  25 , and is without distortion components at frequencies f3 and f4 in the ideal case. 
     While this conventional RF amplifier topology ideally eliminates distortion products, the conventional RF amplifier topology does not provide closed-loop control, manifesting itself by degraded performance caused by changes in component characteristics resulting from, for example, temperature variations and by component aging. What is needed is an RF amplifier topology that provides simple closed-loop control for both loops, thus eliminating performance degradation as component characteristics change with temperature variations and component aging. 
     SUMMARY OF THE INVENTION 
     The present invention provides an RF amplifier topology having closed-loop control and eliminating performance degradation as component characteristics drift caused by temperature variations and component aging. 
     The advantages of the present invention are provided by a multi-tone signal amplifier topology and an amplifying method in which a first amplifier outputs a first signal having at least one fundamental frequency signal and a first distortion signal. A first coupler is coupled to the first signal and outputs a first coupled signal. A phase shifter, coupled to the first coupled signal, outputs a phase-delayed first coupled signal that is phase delayed by substantially a 180° phase delay. A second coupler is coupled to the phase-delayed first coupled signal and outputs at least a portion of an input signal for a second amplifier. 
     The second amplifier outputs a second signal that has a fundamental frequency signal corresponding to each fundamental frequency signal of the first signal and a second distortion signal. Each fundamental frequency signal of the second signal is substantially in-phase with the corresponding fundamental frequency signal of the first signal, while the second distortion signal is substantially 180° out-of-phase with the first distortion signal. An output coupler combines the first and second signals to form a third signal having the corresponding fundamental frequency signals of the first and second signals constructively combined and a third distortion signal that is a difference between the first distortion signal and the second distortion signal. Each fundamental frequency signal of both the first and second signals has a first power level, such that each fundamental frequency signal of the third signal has a power level that is substantially twice the first power level. 
     A detector, coupled to the third signal, generates a control signal applied to the second amplifier for adjusting at least one of an amplitude and a phase of the second distortion signal for minimizing the third distortion signal. The second amplifier circuit includes an adjust circuit that varies at least one of an amplitude and a phase of the input signal for the second amplifier in response to the control signal. 
     According to the invention, the first signal includes a first spread inject signal, and the second signal includes a second spread inject signal. The detector detects the first spread inject signal and the second spread inject signal and generates the control signal based on the detected first spread inject signal and the second spread inject signal. 
    
    
     BRIEF DESCRIPTION OF THE DRAWING 
     The present invention is illustrated by way of example and not limitation in the accompanying figures in which like reference numerals indicate similar elements and in which: 
     FIG. 1 shows a schematic block diagram of a conventional low-distortion RF amplifier topology for linearly amplifying an input signal; and 
     FIG. 2 shows a schematic block diagram of a linear RF amplifier topology having closed-loop control according to the present invention. 
    
    
     DETAILED DESCRIPTION 
     The present invention provides an RF amplifier topology having close-loop control that cancels undesired amplifier intermodulation products that are caused by the nonlinear nature of an RF amplifier, while simultaneously providing the ability to monitor the degree of cancellation of the intermodulation products, that is, the level of non-linearity of the amplifier circuit. Further, the amplifier topology of the present invention provides an adjustment for changes in component characteristics caused by, for example, temperature variations and component aging, thereby obtaining optimal linearity performance of an RF amplifier. 
     FIG. 2 shows a schematic block diagram of a linear RF amplifier topology  30  having closed-loop control according to the present invention. Amplifier topology  30  is described herein for convenience as a low-distortion amplifier for amplifying a high-frequency, dual-tone input signal V in . It is understood, however, that the invention is equally applicable in low-distortion amplification of input signals having more than two tones. Moreover, the present invention may be used for low-distortion amplification of a single tone input signal in situations where the amplifier elements used would otherwise tend to produce undesirable IMD components. 
     To illustrate the advantages of the present invention, linear amplifier equations are formed for each of three loops of circuit  30 . All coefficients in the loop equations are in terms of voltage. For example, for a coupler C, k1 2 +k2 2 =1, if coupler C is lossless, and is less than 1 if losses are considered. 
     Loop 1 has two paths that meet at NODE A: 
     Path 1: CO 2 -C 1 -β 1 -g 1 -C 2 -α 3 -C 3   
     Path 2: CO 2 -C 1 -τ 1 -C 3   
     For Path 1 of loop 1, signal V in  passes through the direct path of a coupler CO 2  and a coupler C 1 . Path 1 of loop 1 then includes an attenuation and phase adjust circuit β 1 , an amplifier g 1 , a coupler C 2 , an attenuator α 3  and a coupling port of a coupler C 3  before arriving at NODE A. Path 2 of Loop 1 includes the direct port of coupler CO 2 , the coupling port of coupler C 1 , a delay line τ 1 , and the direct port of coupler C 3  before arriving at NODE A. 
     A spread signal INJECT  1  is input into path 1 of Loop 1 through the coupled port of a coupler CDI 1 . Spread signal INJECT  1  and a spread signal INJECT  2 , which is input through a coupler CDI 2  and is described below, are used for controlling loop 2. For analysis purposes, the amplitude and phase contributions of the through-path of couplers CDI 1  and CDI 2  are absorbed into the amplitude and phase adjust circuits β 1 e jψ1  and β 2 e jψ2 , respectively. 
     The loop equation for path 1 of loop 1 is: 
      ( k 10 e   jφ10 )( k 1V in   e   jφ1 )(β 1   e   jψ1 )(g 1   e   jθ1 )( k 5 e   jφ5 )(α 3   e   jτ3 )( k 6 e   jφ6 )  } Main Term 
     
       
          +δ1( k 10 e   jφ10 )( k 1V in   e   jφ1 )(β 1   e   jψ1 )(g 1   e   jθ1 )( k 5 e   jφ5 )(α 3   e   jτ3 )( k 6 e   jφ6 )  } Dist. Term (1) 
       
     
     where, the Main Term is the desired signal and Dist. Term is the distortion term. 
     The loop equation for loop 1, path 2 is: 
     
       
         ( k 10 e   jφ10 )( k 2V in   e   jφ2 )(α 1   e   jτ1 )( k 4 e   jφ4 )  } Main Term (2) 
       
     
     The sum of the main terms of Eqs. (1) and (2) is set equal to (k10e jφ10 )(k1V in e jφ1 ) at NODE A so that the distortion term of the signal input to amplifier g 3 , just prior to amplitude and phase adjust circuit β 2 e jψ1 , equals the main term of the signal input to amplifier g 1 , just prior to amplitude and phase adjust circuit β 1 e jψ1 . If amplifier g 1  and amplifier g 3  are identical and β 1  and β 2  are similarly identical, the same distortion terms will be produced by amplifier g 1  and by amplifier g 3 . To ensure that the sum of the main terms of the signals appearing at NODE A equals (k10e jφ10 )(k1V in e jφ1 ), signal DIFF.DET. 1  and DIFF.DET. 2  are input to couplers CDI 1  and CDI 2 , respectively, for adjusting for any difference between the sum of the main terms and (k10e jφ10 )(k1V in e jφ1 ). 
     Thus, at NODE A, 
     
       
         V NODE A =( k 10 e   jφ10 )( k 1V in   e   jφ1 )(β 1   e   jψ1 )(g 1   e   jθ1 )( k 5 e   jφ5 )(α 3   e   jτ3 )( k 6 e   jφ6 )  } Main Term (Path 1) 
       
     
     
       
          +( k 10 e   jφ10 )( k 2V in   e   jφ2 )(α 1   e   jτ1 )( k 4 e   jφ4 )  } Main Term (Path 2) 
       
     
     
       
          +δ1( k 10 e   jφ10 )( k 1V in   3   e   jφ1 )(β 1   e   jψ1 )(g 1   e   jθ1 )( k 1V in   e   jφ5 )(α 3   e   jτ3 )( k 6 e   jφ6 )  } Dist. Term (3) 
       
     
     By setting the sum of the main terms of Eq. (3) equal to (k10e jφ10 )(k1V in e jφ1 ), the (k10e jφ10 ) terms cancel, resulting in: 
     
       
           k 1V in   e   jφ1 =V in [( k 1β 1 g 1   k 5α 3   k 6) e   j(φ1+ψ1+θ1+φ5+τ3+φ6) +( k 2α 1   k 4) e   j(φ2+τ1+θ4) ]  } Main Term (4) 
       
     
     The phase of the distortion term generated by g 1 , after passing through amplitude and phase adjust circuit β 2 e jψ2  and through amplifier g 3 , should be 180° out-of-phase with the distortion term generated by amplifier g 3  by amplifying the main term k1V in e jθ1  so that the distortion terms at the output of amplifier g 3  cancel. Thus, 
     
       
         φ1+ψ2+θ3=φ1+ψ1+θ1+φ5+τ 3 +φ6+ψ2+θ3+180°   (5) 
       
     
     Rewriting Eq. (5), 
     
       
         −180°=φ5+τ 3 +φ6+ψ1+θ1  (6) 
       
     
     Substituting Eq. (6) into Eq. (4) yields, 
     
       
           k 1V in   e   jφ1 =V in [(− k 1β 1 g 1   k 5α 3   k 6) e   j(φ1) +( k 2α 1   k 4) e   j(φ2+τ1+φ4) ]  (7) 
       
     
     Setting the phase terms of Eq. (7) equal to each other, 
     
       
         φ1=φ1=φ2+τ 1 +φ4  (8) 
       
     
     Solving Eq. (7) for the amplitude terms, 
     
       
           k 1=( k 2α 1   k 4)−( k 1β 1 g 1   k 5α 3   k 6)  (9) 
       
     
     Rewriting Eq. (9), 
     
       
         ( k 2α 1   k 4)= k 1(1+β 1 g 1   k 5α 3   k 6)  (10) 
       
     
     Using Eqs. (4), (6), (8) and (9), and rewriting Eq. (3), the voltage at NODE A is: 
     
       
         V NODEA   =k 1 k 10V in   e   jφ1   e   jφ10 −δ1V 3   in   k 1β 1 g 1   k 5 k 10α 3   k 6 e   jφ1   e   jφ10   (11) 
       
     
     The voltage at NODE A is ˜k1k10V in e jφ1 e jφ10  because the distortion term (V in   3  term) is small in comparison to the main term (V in  term). 
     Ideally, the main voltage terms add in-phase at the output of circuit  30  and the distortion terms cancel. 
     Turning now to loop 2, there are two paths that meet at the output V o : 
     Path 1: C 2 -τ 2 -C 4   
     Path 2: β 2 -g 3 -C 4   
     For path 1 of loop 2, input signal V in  passes through the direct port of both couplers CO 2  and C 1 . Path 1 of loop 2 then passes through attenuation and phase adjust circuit β 1 , amplifier g 1 , a direct port of coupler C 2 , a delay line τ 2 , a direct port of C 4 , and a direct port of coupler CO 1  before arriving at V o . Path 2 of loop 2 starts at NODE A and continues through the direct port of coupler CDI 2 , through an attenuation and phase adjust circuit β 2 , an amplifier g 3 , a coupling port of coupler C 4 , and the direct port of coupler CO 1  before arriving at V o . The amplitude and phase contribution of coupler CO 1  is ignored because coupler CO 1  contributes the same term to both paths 1 and 2 of loop 2. 
     The loop equation for path 1 of loop 2 is: 
     
       
         ( k 10 e   jφ10 )( k 1V in   e   jφ1 )(β 1   e   jψ1 )(g 1   e   jθ1 )( k 3 e   jφ3 )(α 2   e   jτ2 )( k 8 e   jφ8 )  } Main Term 
       
     
     
       
          +δ1( k 10 e   jφ10 )( k 1V in   3   e   jφ1 )(β 1   e   jψ1 )(g 1   e   jθ1 )( k 3 e   jφ3 )(α 2   e   jτ2 )( k 8 e   jφ8 )  } Dist. Term (12) 
       
     
     The loop equation for path 2 of loop 2 is: 
     
       
         ( k 10 e   jφ10 )( k 1V in   e   jφ1 )(β 2   e   jψ1 )(g 3   e   jθ3 )( k 7 e   jφ7 )  } Main Term 
       
     
     
       
          −δ1V in   3 ( k 1β 1 g 1   k 5α 3   k 6β 2 g 3   k 7 k 10) e   j(φ+ψ2+θ3+φ7+φ10)   } ‘g 1 ’ Distortion Term 
       
     
     
       
          +δ3V in   3 ( k 1 k 10β 2 g 3   k 7) e   j(φ1+ψ2+θ3+φ7+φ10)   } ‘g 3 ’ Distortion Term (13) 
       
     
     At output V o , the main terms add in-phase. Therefore, 
     
       
         φ10+φ1+ψ1+θ1+φ3+τ 2 +φ8=φ10+φ1+ψ2+θ3+φ7  (14) 
       
     
     Rewriting Eq. (14), 
     
       
         ψ1+θ1+φ3+τ 2 +φ8=ψ2+θ3+φ7  (15) 
       
     
     When g 2  and g 3  are identical and ψ1=ψ2, φ3+τ 2 +φ8=φ7. For the distortion terms to cancel, the amplitude terms must be equal and the phase terms must be equal because the distortion term in Eq. (13) already had a negative sign. From Eqs. (12) and (13), setting the phase of the distortion terms to be equal results in: 
     
       
         φ10+φ1+ψ1+θ1+φ3+τ 2 +φ8=φ10+φ1+ψ2+θ3+φ7  (16) 
       
     
     Note, however, that Eq. (16) is the same as Eq. (14). From Eqs. (12) and (13), setting the amplitude of the distortion terms to sum to zero results in: 
     
       
         δ1 k 1 k 10β 1 g 1   k 3α 2   k 8+δ3 k 1 k 10β 2 g 3   k 7−δ1 k 1 k 10β 1 g 1   k 5α 3   k 6β 2 g 3   k 7=0  (17) 
       
     
     When g 1  and g 3  are identical and β 1 =β 2 , k3α 2 k8+k7=k5α 3 k6β 2 g 3 k7. 
     Finally, at the output V o , the main term is: 
     
       
         Vo=( k 10 e   jφ10 ) k 1V in   e   jφ1 (β 1   e   jψ1 )(g 1   e   jθ1 )( k 3 e   jφ3 )(α 2   e   jτ2 )( k 8 e   jφ8 ) 
       
     
     
       
          + k 1V in   e   jφ1 ( k 10 e   jφ10 )(β 2   e   jψ2 )(g 3   e   jφ3 ) k 7 e   jφ7 )  } Main Term (18) 
       
     
     Rewriting Eq. (18) by using Eq. (16) results in: 
     
       
         Vo= k 1V in   e   jφ1   k 10 e   jφ10 (β 1 g 1   k 3α 2   k 8+β 2 g 3   k 7) e   j(ψ1+θ1+φ3+τ2+φ8)   (19) 
       
     
     The third loop is used for monitoring and cancelling any distortion term present at V o  that is caused by changes in component characteristics resulting from, for example, temperature variations and/or by component aging. To do this, the main terms of V in  and V o  are subtracted at a monitor port NODE B and a broadband diode detector D 1  is used for detecting the remaining distortion term present at NODE B. An attenuation and phase adjust circuit β 3  is then adjusted based on digital processing and control so that the main terms will cancel more effectively. 
     Once the main terms have been canceled, the remaining distortion signal, of which the INJECT  1  and INJECT  2  signals are part, is despread and detected by a narrowband detector. At this point, the detected spread tone is canceled by controlling attenuation and phase adjust circuit β 2  of loop 2. By canceling, or minimizing, the spread tones, any distortion term at V o  is also minimized. 
     Loop 3 has two paths that meet at NODE B: 
     Path 1: CO 1 -CO 3   
     Path 2: CO 2 -β 3 -CO 3 -τ 4   
     Path 1 of loop 3 passes through the coupling ports of coupler CO 1  and a coupler CO 3 . Path 2 of loop 3 passes from the input through the coupling port of coupler CO 2 , through a delay line α 4 , through an attenuation and phase adjust circuit β 3 , and finally through the direct port of coupler CO 3 . The input signal to coupler CO 1  is expressed in Eq. (19) as: 
     
       
         Vo= k 1V in   e   jφ1   k 10 e   jφ10 (β 1 g 1   k 3α 2   k 8+β 2 g 3   k 7) e   j(ψ1+θ1+φ3+τ2+φ8)   (20) 
       
     
     The loop equation for path 1 of loop 3 is: 
     
       
         ( k 1V in   e   jφ1   k 10 e   jφ10 (β 1 g 1   k 3α 2   k 8+β 2 g 3   k 7) e   j(ψ1+θ1+φ3+τ2+φ8) )( k 15 e   jφ7 )( k 11 e   jφ11 )  (21) 
       
     
     The loop equation for path 2 of loop 3 is: 
     
       
         ( k 9V in   e   jφ9 )(β 3   e   jψ3 )( k 12 e   jφ12 )(α 4   e   jτ4 )  (22) 
       
     
     Summing Eqs. (20) and (21) results in: 
     
       
         V NODE B =( k 1V in   e   jφ1 (β 1 g 1   k 3α 2   k 8+β 2 g 3   k 7) e   j(ψ1+θ1+φ3+τ2+φ8) )( k 15 e   jφ7 )( k 11 e   jφ11 ) 
       
     
     
       
          +( k 9V in   e   jφ9 )(β 3   e   jψ3 )( k 12 e   jφ12 )(α 4   e   jτ4 )  (23) 
       
     
     The main terms in Eq. (23) are set to be equal in amplitude and cancel in phase by adjusting β 3 e jψ3 . 
     While the present invention has been described in connection with the illustrated embodiments, it will be appreciated and understood that modifications may be made without departing from the true spirit and scope of the invention.