Abstract:
A method for reducing effects of terrain return fading due to summation of out of phase radar returns in determining locations of radar targets is described. The method comprises determining an interferometric angle, Φ, to a radar target based on at least one radar return and filtering the interferometric angle, Φ, by adjusting an effect of terrain features contributing to the interferometric angle, Φ, proportionally to a degree of radar return fading resulting from the terrain features of the radar targets. A corrected interferometric angle, Φ out ., is then provided, based at least in part on the filtering.

Description:
BACKGROUND OF THE INVENTION 
     This invention relates generally to radar systems, and more specifically to a radar system which is capable of synchronization with a digital elevation map (DEM) to accurately determine a location. 
     The proper navigation of an aircraft in all phases of its flight is based to a large extent upon the ability to determine the terrain and position over which the aircraft is passing. In this regard, radar systems, altimeters, and other instrumentation in combination with the use of accurate electronic terrain maps, aid in determining the flight path of the aircraft. Electronic terrain maps provide the height of objects on a map and are presently used to assist in the navigation of aircraft. 
     Pulse radar altimeters accurately determine altitude using leading edge return signal tracking. Specifically, a pulse radar altimeter transmits a pulse of radio frequency (RF) energy, and a return echo is received and tracked using a tracking system. The interval of time between signal bursts of a radar system is referred to as a pulse repetition interval (PRI). The frequency of bursts is referred to as a pulse repetition frequency (PRF) and is the reciprocal of PRI. 
     A radar altimeter mounted on an aircraft transmits a signal that impinges a ground patch bounded by an antenna beam. As is well known, the Doppler effect results in isodops based on selection by Doppler filters within the radar altimeter. The area between the isodops is referred to as swaths. The Doppler filter, and resulting isodops are well known in this area of technology and will not be explained in any further detail. To scan a particular area, range gates are used to further partition the swath created by the Doppler filter. To scan a certain Doppler swath, many radar range gates operate in parallel. With the range to each partitioned area determined, a record is generated representing the contour of the terrain below the flight path. The electronic maps are used with the contour recording to determine the aircraft&#39;s position on the electronic map. This system is extremely complex with all the components involved as well as the number of multiple range gates that are required to cover a terrain area. As a result, the computations required for this system are very extensive. 
     In addition to the complexity, the precision and accuracy of the distance to a particular ground area or object has never been attained using an airborne radar processor. 
     BRIEF SUMMARY OF THE INVENTION 
     In one aspect, a method for determining location of a radar target is provided. The method comprises determining an interferometric angle, Φ, to the radar target based on at least one radar return and filtering the interferometric angle, Φ, through an adjustment of the effect of terrain features contributing to the interferometric angle, Φ. The adjustment is proportional to a degree of radar return fading resulting from the terrain features of the radar target. A corrected interferometric angle, Φ out , is then provided, based at least in part on the filtering. 
     In another aspect, an apparatus for filtering an interferometric angle to a radar target to counter effects of terrain return fading due to summation of out of phase returns in the calculation of the interferometric angle to the radar target is provided. The apparatus comprises at least one signal fade detector, each detector receiving a signal representative of a radar return for one of a plurality of radar channels and outputting a quality factor signal, Q fade , that represents a measure of a depth of the signal fade for that radar channel. The apparatus also comprises a filter which receives the interferometric angle, Φ in (n), and the quality factor signals. The filter outputs a weighted interferometric angle, Φ out (n), which is weighted according to the received quality factor signals. 
     In still another aspect, a unit for determining an interferometric angle to a radar target is provided. The unit receives radar return data from a right radar channel, a left radar channel, and an ambiguous radar channel, where the ambiguous radar channel has an antenna located between antennas for the left and right radar channels. The unit comprises a phase processor receiving the radar return data from the right, left, and ambiguous radar channels. The phase processor determines the phase between the left radar channel and ambiguous radar channel return data, the phase between the ambiguous radar channel and right radar channel return data, and the phase between the right radar channel and left radar channel return data. The unit also comprises a phase ambiguity processor receiving the determined phases from the phase processor, and calculating a preliminary interferometric angle, Φ in (n), to the radar target. The unit further comprises a plurality of signal fade detectors and a filter. One detector receives right radar channel return data, a second detector receives left radar channel return data, and a third detector receives ambiguous radar channel return data. Each detector outputs a quality factor signal that represents a measure of a depth of the signal fade for that radar channel. The filter receives the preliminary interferometric angle, Φ in (n), from the phase ambiguity processor and the quality factor signals from the signal fade detectors. The filter outputs a weighted interferometric angle, Φ out (n), which is weighted according to the received quality factor signals. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 a  is a diagram illustrating swaths made by a radar. 
     FIG. 1 b  is a diagram illustrating a radar transmit pattern. 
     FIG. 2 is an illustration of radar signal waveforms over time. 
     FIG. 3 is a diagram illustrating radar signals being received by three antennas. 
     FIG. 4 is a diagram illustrating a body coordinate system. 
     FIG. 5 is a diagram illustrating a doppler coordinate system with respect to the body coordinate system of FIG.  4 . 
     FIG. 6 is a block diagram of a radar signal processing system. 
     FIG. 7 is a block diagram of a digital sampling and filtering section. 
     FIG. 8 is a block diagram of a correlation band pass filter. 
     FIG. 9 is a block diagram of a in-phase/quadrature mixer. 
     FIG. 10 is a block diagram of an all pass filter network for in-phase and quadrature components of a signal, within the mixer of FIG.  8 . 
     FIG. 11 is a diagram of a second order all pass filter. 
     FIG. 12 is a block diagram of a swath band pass filter. 
     FIG. 13 is a block diagram of a filter coefficients processor. 
     FIG. 14 is a velocity vector diagram. 
     FIG. 15 is a block diagram of a phase processor including three phase detectors. 
     FIG. 16 is a block diagram of one phase detector from FIG.  15 . 
     FIG. 17 is a block diagram of an interferometric angle resolver. 
     FIG. 18 is a chart illustrating varying electrical phase differences between three antenna pairings. 
     FIG. 19 is a block diagram which illustrates inputs to a body coordinate processor. 
     FIG. 20 is a block diagram of the body coordinate processor of FIG.  19 . 
     FIG. 21 is an illustration of the derivation of a doppler circle. 
     FIG. 22 is an illustration of the derivation of an interferometric circle. 
     FIG. 23 is an illustration of a unit which applies filtering to an interferometric angle. 
     FIG. 24 is a detailed illustration of one embodiment of signal fade detectors. 
     FIG. 25 is a diagram illustrating barker coded transmit and receive pulses. 
     FIG. 26 is a block diagram illustrating inputs to and outputs from a range verification processor. 
     FIG. 27 is a flowchart illustrating a range verification method. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 1 shows an aircraft  2  with the Doppler effect illustrated by isodops as a result of selection by the use of Doppler filters. For ease of description aircraft  2  is assumed to have a vertical velocity of zero. As is known, if a vertical velocity exists, the median  8  of the Doppler effect will shift depending on the vertical velocity. If the aircraft  2  has a vertical velocity in a downward direction, the median of the Doppler would shift to the right of the figure. If the aircraft  2  has a vertical velocity in an upward direction, the Doppler would shift to the left of the figure. 
     There is herein described a combination Doppler radar/interferometer to navigate an aircraft  2  with respect to terrain features below aircraft  2 . As used herein, aircraft is used to identify all flight platforms which may incorporate a radar system, including, but not limited to, jets, airplanes, unmanned aerial vehicles, missiles, and guided weapons. The radar also functions with an electronic map, sometimes referred to herein as a digital elevation map (DEM), in determining a position of aircraft  2 . In addition to determining an altitude of aircraft  2 , an XYZ location of the nearest object to aircraft  2  on the ground, with respect to aircraft  2  in a certain terrain area can be determined. As aircraft  2  is flying over terrain as shown in FIGS. 1 a  and  1   b , it is important to determine a position of aircraft  2  in accordance with a map. A Doppler filter and range gate are used with a transmitted beam  10  from a transmit antenna (not shown in FIG.  1 ). 
     In a general altitude range tracking radar (e.g. radar altimeter), range is measured and indicated by measuring the time for transmitted energy to be reflected from the surface and returned. With reference to FIG. 2, a radar transmitter repeatedly sends out bursts of electromagnetic energy at a predetermined repetition rate from an antenna, as indicated by transmit pulse  20 . Following a time delay which is a function of the aircraft altitude, a ground return pulse  22  is received by a receiving antenna feeding a receiver. A range gate  30  is utilized by the tracking radar to view at least a portion of ground return  22 . 
     Referring to FIG. 3, three receive antennas, a right antenna (R)  42 , a left antenna (L)  44 , and an ambiguous antenna (Amb)  46 , are used to receive information. Along with the three antennas, three processing channels, referred to below as left, right and ambiguous respectively, each include a receiver, a data acquisition device, range gate, and a filter. Use of the three antenna system, along with the processing described herein, provides a solution to ambiguous detected angle of the nearest object. The ambiguous detected angle is due to the spacing of the antennas being greater than the transmitted RF frequency wavelength. By receiving three returns, the processing system is able to determine the unambiguous angle to the nearest object on the ground, which in turn is utilized to locate position of aircraft  2  in body coordinates. Body coordinates are typically preferable than positioning as determined by known systems, as those systems determine position as if the body aircraft  2  is aligned with the line of flight. As aircraft  2  is prone to pitch, roll, and yaw, the body of aircraft  2  is not necessarily aligned with the line of flight. 
     In an exemplary illustration, right antenna  42 , along with processing systems (described below) will provide a course range search which roughly determines the range to the nearest point  48  in swath  12  (shown in FIG. 1) before aircraft  2  has passed over from swath  14  into swath  12 . Determination of the nearest point  48  is performed by a wide bandwidth, high speed track loop which quickly determines the range to nearest point  48  in swath area  12 . Nearest point  48  provides a starting point for a tracking loop using left antenna  44  and ambiguous antenna  46 . The track loop controls the range gate to track returns from a transmit antenna. A narrow bandwidth, high precision processor is used to set range gates for left antenna  44  and ambiguous antenna  46  to an exact range of nearest point  48  based on the previous course range determination. The operation of the three receive antennas and associated processing channels provides a quick and accurate setting of a range gate on the nearest object in the Doppler swath  14  directly below aircraft  2  so that a phase difference can be measured and along with the known separations  50  amongst the three antennas, a crosstrack distance to the object  48  is determined. The crosstrack distance is the distance, horizontal and perpendicular to the body coordinates of aircraft  2 , to object  48 . 
     FIG. 3 shows a view with aircraft  2  going into the Figure. During the phase comparison portion of the time interval, the Doppler filters of the left, right and ambiguous channels are set to select a swath  14  (shown in FIG. 1) below aircraft  2 . Further, both range gates are set at a range directly on the nearest object  48  as previously determined. From this range, right antenna  42  receives a signal from object  48  at a distance of R 1 , ambiguous antenna  46  receives a signal from the object  48  at a distance of RA, and left antenna  44  receives the signal from object  48  at a distance of R 2  where the distance difference is a function of the antenna separation  50  between and amongst the three antennas. A phase processor (described below) compares the phase difference between R 1  and RA, R 2  and RA, and R 1  and R 2  once the return signals are received. As illustrated in the Figure, the exact range differences (R 2 −R 1 ), (RA−R 1 ), and (R 2 −RA) are from phase differences and simple trigonometry relations are used to determine the exact crosstrack distance to the object  48  in aircraft body coordinates. 
     As illustrated in FIG. 3, after the range differences (R 2 −R 1 ), (RA−R 1 ), and (R 2 −RA) are determined and knowing the antenna separations  50 , and measured range R 1 , then the crosstrack distance (Y) and vertical distance (Z) can also be computed in aircraft body coordinates. It is important that the precise location of nearest object  48  in each swath is determined so correlation can be made with the electronic maps which will accurately locate the aircraft  2  on the electronic map. For example, at typical high speed aircraft cruising velocities, a radar, configured with reasonably sized Doppler filters, has swath widths of approximately 10 feet at 5000 feet altitude. The resulting incidence angle formed by the intersection of R 1  and a vertical line  27  will then be on the order of less than 3 degrees. Basic trigonometry relations show that even with a typical error (for example 1%) on the radar range gate measured distance R 1 , (50 feet at 5000 feet altitude), knowing the precise antenna separation  50 , and precise range differences (R 2 −R 1 ), (RA−R 1 ), and (R 2 −RA), the crosstrack distance (Y) will be precise due to the very small incidence angle encountered. 
     FIG. 4 illustrates a body coordinate system. The body coordinate system, is the coordinate system with respect to aircraft body  2 . An x-axis, Xm is an axis which passes through a nose of aircraft body  2 . A y-axis, Ym, is an axis which is 90 degrees from Xm and is positive to the right of aircraft body  2 . A z-axis, Zm, is an axis which is 90 degrees from both Xm and Ym and perpendicular to a bottom of aircraft body  2 . With respect to aircraft maneuvering, a positive roll is a drop of the right wing, a positive pitch is a nose up, and a positive yaw is the nose to the right, all with respect to a line of flight. 
     It is known that aircraft do not typically fly in alignment with the aircraft body coordinates. Such a flight path is sometimes referred to as a line of flight. Therefore an aircraft which is flying with one or more of a pitch, roll, or yaw, and which has a hard mounted radar system, introduces an error element in a determination of target location, in body coordinates. As such radars typically operate with respect to the line of flight, a coordinate system with respect to the line of flight has been developed and is sometimes referred to as a doppler coordinate system. FIG. 5 illustrates differences between aircraft coordinates and doppler coordinates. An x-axis of the doppler coordinate system, Xd, is on the line of flight. A y-axis, Yd, and a z-axis, Zd, at right angles to Xd, respectively are defined as across Xd, and above and below Xd. 
     Therefore, if aircraft  2  is flying with no pitch, roll, or yaw, the body coordinate system aligns with the doppler coordinate system. For a positive roll, Xm and Xd are still aligned, while Yd rotates below Ym and Zd rotates to the left of Zm. For a positive yaw, Xd rotates to the right of Xm, Yd rotates behind Ym, and Zd and Zm are aligned. For a positive pitch, Xd rotates above Xm, Yd aligns with Ym, and Zd rotates ahead of Zm. The complexity of having multiple of pitch, roll, and yaw, and determining a target position in aircraft body coordinates is apparent. 
     FIG. 6 is one embodiment of a doppler radar processing system  200 . System  200  incorporates three radar antennas which receive reflected radar pulses, the pulses having originated from a radar source. A left antenna  202  receives the pulses and forwards the electrical signal to receiver  204 . Receiver  204  forwards the received radar signal to a data acquisition unit  206 . A right antenna  208  receives the pulses, at a slightly different time than left antenna  202 , and forwards the electrical signal to receiver  210 . Receiver  210  forwards the received radar signal to a data acquisition unit  212 . An ambiguity antenna  214  also receives the reflected radar signal, and passes the received signal to a circulator  216 . Circulator  216  functions to direct the transmit signal to the antenna, and to direct the received signal from the antenna to receiver  220 , thereby allowing a single antenna to be used for both transmitting and receiving. Receiver  220  forwards the received signal to a data acquisition unit  222 . 
     Data acquisition unit  206  provides a digital signal representative of the signal received at left antenna  202  to a left phase pre-processing unit  224 . Similarly, representative signals are received at pre-processing units  226  and  228  from data acquisition units  222  and  212 , respectively. Data acquisition units  206 ,  212 , and  222  are configured, in one embodiment, to sample received signals, and thereby reduce the data to a rate which allows a relatively low speed computer to process digitized radar data. In one embodiment, pre-processing units  224 ,  226 , and  228  perform a gate ranging function. 
     A phase processor  230  receives gated, filtered signals, representative of left, right, and ambiguity signals received at the antennas, and determines a phase relationship between each of the left and ambiguous signal, the right and ambiguous signals, and the right and left signals. The phase relationships between the signals are used, along with slant range, velocity and attitude readings in a phase ambiguity processing unit  232  to determine an interferometric angle, Φ, to a target. A body coordinate processor  233  utilizes the interferometric angle, Φ, to determine an XYZ position of, for example, an aircraft employing system  200  with respect to a current aircraft position, sometimes referred to herein as aircraft body coordinates. 
     A signal from data acquisition unit  222  is also received at an automatic gain control (AGC) unit  234 . A signal from AGC unit  234  is passed to pre-processing units  236 ,  238 , and  240 . A filtered signal from pre-processing unit  236  is passed to range track processor  242  which provides a slant range signal to phase ambiguity processing unit  232  and altitude information. Pre-processing unit  238  passes a filtered signal to a range verification processor  244 . Pre-processing unit  240  passes a filtered signal to a range level processor  246 , which also provides a feedback signal to AGC  234 . 
     FIG. 7 is a block diagram of a digital processing section  300  for system  200  (shown in FIG.  6 ). Components in section  300 , identical to components of system  200 , are identified in FIG. 7 using the same reference numerals as used in FIG.  6 . Section  300  includes pre-processing units  224 ,  226 ,  228 ,  236 ,  238 , and  240  and processors  230 ,  242 ,  244 , and  246 . Referring specifically to pre-processing units  224 ,  226 ,  228 ,  236 ,  238 , and  240 , each includes a gate correlator  302 , a correlation band pass filter  304 , a baseband I/Q mixer  306 , and a swath band pass filter  308 . A filter coefficients processor  309 , in one embodiment, is configured to provide at least a filter center frequency in hertz, Fc, a filter bandwidth in hertz, B, and a filter sampling frequency in hertz, Fs, to swath band pass filter  308 , which uses Fc, B, and Fs in determination of filter coefficients. In one embodiment, processor  309  receives as input, an antenna mounting angle, velocity vectors in body coordinates, a pitch, and a slant range. 
     FIG. 8 is a block diagram of a correlation band pass filter  304  (also shown in FIG.  7 ). An input signal  310 , sometimes referred to as x(0), is fed into a summing element  312 . An output of summing element  312  is multiplied by a coefficient  313 , which, in one embodiment has a value of 1/K 1  (further described below). After multiplication by coefficient  313 , an output signal  314 , sometimes referred to as y(0), is generated. Another input into summing element  312  is provided by input signal  310  being delayed by a two sample delay element  316 , whose output, sometimes referred to as x(−2), is fed into summing element  312 . Further, output signal  314  is fed back into a second two sample delay element  318 , whose output, sometimes referred to as y(−2), is multiplied by a second coefficient  319 , and fed into summing element  312 . In one embodiment, coefficient  319  has a value of K 3 . Therefore, a present output, y(0) is calculated as y(0)=(1/K 1 )×[x(0)−x(−2)]−(K 2 ×y(−2)), where K 1 =C+1, K 3 =C−1, K 2 =K 3 /K 1 , and C=1/Tan(π×bandwidth/f sample ) where bandwidth and sample frequency are in hertz, and the angle for which the tangent is to be calculated is in radians. 
     In alternative embodiments, filter  304  is configured to filter range ambiguity spectrum lines, filter out-of-band interference signals and stretch the input signal, which is a pulse, to a continuous wave (CW) signal. Filter  304 , in one embodiment, receives as input an output of gate/correlator  302  (shown in FIG. 7) at a sample rate of 100 MHz, an IF frequency of 25 MHz, and has a bandwidth of 10 KHz. Therefore, in this embodiment, there are four samples per IF frequency period. 
     A sample clock at 100 MHz provides samples at a 10 nsec rate. For example, a 4 μsec pulse repetition interval (PRI) (N=400 clocks per PRI) and two sample gate width, results in two non-zero gated return samples, x(0) and x(1), and  398  zero amplitude samples, x(2)−x(399), into correlation filter  304  during one PRI. In order to provide a filter of reasonable processing size and speed, the zero amplitude samples which do not affect filter output are not processed by filter  304 . Therefore, past outputs, for example y(−2), required in the filter feedback configuration, as illustrated by delay elements  316  and  318 , at the time of non-zero inputs are not available. These past outputs are calculated based on filter outputs generated during and directly after the previous return (the previous non-zero samples), and filter droop characteristics over a known pulse repetition interval. 
     In addition, one of the past outputs, y(−1), is not used because it has a feedback multiplier with a value of nearly zero in one embodiment of filter  304 , because of the narrow 10 kHz bandwidth. 
     In one exemplary embodiment, where F sample =100 MHz, center frequency=25 MHz, and Bandwidth=8 KHz, coefficients are calculated as K 1 =3979.873661, K 3 =3977.873661, and K 2 =0.9994974715. Let P=the number of samples in a PRI. Filter  304  starts calculating at the beginning of a gate width and continues for two counts after the end of the gate width. After the gate width +2 counts the next step is to calculate y(−2) and y(−1) and wait for x(P) data, the beginning of the next gate width, where x(P) is equivalent to x(0). Table 1 illustrates a general procedure for operation of filter  304 , for low altitude radar data, track and phase gate of two sample widths, and a PRI of 400 μsec. The calculation for filter output y(0) requires filter output y(−2). The example of Table 2 example illustrates calculation of y(−2) where N=400, if PRI=4 μsec. 
     
       
         
               
             
               
               
               
             
               
               
               
             
           
               
                 TABLE 1 
               
             
             
               
                   
               
               
                 Correlation Filter Algorithm Example 
               
             
          
           
               
                 x(N) 
                 Count (N) 
                 Algorithm 
               
               
                   
               
             
          
           
               
                 0 
                 397 
                 y(−3) = y(397) 
               
               
                 0 
                 398 
                 y(−2) = y(398) 
               
               
                 0 
                 399 
                 y(−1) = y(399) 
               
               
                 x(0) 
                 0 
                 y(0) = (1 / K1)[x(0) − x(−2)] − [K2 x y(−2)] 
               
               
                 x(1) 
                 1 
                 y(1) = (1 / K1)[x(1) − x(−1)] − [K2 x y(−1)] 
               
               
                 0 
                 2 
                 y(2) = (1 / K1)[x(2) − x(0)] − [K2 x y(0)] 
               
               
                 0 
                 3 
                 y(3) = (1 / K1)[x(3) − x(1)] − [K2 x y(1)] 
               
               
                 0 
                 4 
                 y(4) = 0 − K2xy(2) = −K2xy(2) = (−K2) 1 xy(2) 
               
               
                 0 
                 5 
                 y(5) = 0 − K2xy(3) = −K2xy(3) = (−K2) 1 xy(3) 
               
               
                 0 
                 6 
                 y(6) = 0 − K2xy(4) = −K2xy(4) = −K2[(−K2)xy(2)] = 
               
               
                   
                   
                 (−K2) 2 xy(2) 
               
               
                 0 
                 7 
                 y(7) = 0 − K2xy(5) = −K2xy(5) = −K2[(−K2)xy(3)] = 
               
               
                   
                   
                 (−K2) 2 xy(3) 
               
               
                 0 
                 8 
                 y(8) = 0 − K2xy(6) = −K2xy(6) = −K2[(−K2)x(−K2) 
               
               
                   
                   
                 xy(2)] = (−K2) 3 xy(2) 
               
               
                 0 
                 9 
                 y(9) = 0 − K2xy(7) = −K2xy(7) = −K2[(−K2)x(−K2) 
               
               
                   
                   
                 xy(3)] = (−K2) 3 xy(3) 
               
               
                 0 
                 10 
                 y(10) = 0 − K2xy(8) = −K2xy(8) = −K2[(−K2) 
               
               
                   
                   
                 x(−K2)x(−K2)xy(2)] = (−K2) 4 xy(2) 
               
               
                 0 
                 11 
                 y(11) = 0 − K2xy(9) = −K2xy(9) = −K2[(−K2) 
               
               
                   
                   
                 x(−K2)x(−K2)xy(3)] = (−K2) 4 xy(3) 
               
               
                   
               
             
          
         
       
     
     In one embodiment, y(399) becomes y(0) if a range gate is moved in an inbound direction. The resulting P becomes 399. If a range gate is moved in an outbound direction, y(1) becomes y(0), and the resulting P becomes 401. Algorithms shown for determination of y(4) through y(11) are used to formulate a general algorithm equation. 
     In addition to an example illustration of calculation of y(−2) with a P of 400 and a gate width of two clock counts, Table 2 also illustrates a general algorithm equation for counts (N) greater than three, (i.e. y(N)=(−K 2 ) M ×y(2), for N even and y(N+1)=(−K 2 ) M ×y(3), where M=(N(even)/2)−1. 
     
       
         
               
             
               
               
               
             
               
               
               
             
           
               
                 TABLE 2 
               
             
             
               
                   
               
               
                 General Algorithm Equation after N = 3 
               
             
          
           
               
                 Bin 
                 Count (N) 
                 Algorithm 
               
               
                   
               
             
          
           
               
                 0 
                 396 
                 y(−4) = (−K2) 197  x y(2) 
               
               
                 0 
                 397 
                 y(−3) = (−K2) 197  x y(3) 
               
               
                 0 
                 398 
                 y(−2) = (−K2) 198  x y(2) 
               
               
                 0 
                 399 
                 y(−1) = (−K2) 198  x y(3) 
               
               
                 x(0) 
                 0 
                 y(0) = (1 / K1)[x(0) − x(−2)] − [K2 x y (−2)] 
               
               
                 x(1) 
                 1 
                 y(1) = (1 / K1)[x(1) − x(−2)] − [K2 x y (−1)] 
               
               
                 0 
                 2 
                 y(2) = (1 / K1)[x(2) − x(0)] − [K2 x y (0)] 
               
               
                 0 
                 3 
                 y(3) = (1 / K1)[x(3) − x(1)] − [K2 x y (1)] 
               
               
                   
               
             
          
         
       
     
     In the embodiment described, for y(0) through y(3), the filter algorithm is because new x(N) and/or y(N) data are available. After the y(3) algorithm calculation, y(398) and y(399) are calculated, and the filter algorithm is configured to wait for x(400) data, where x(400) is equivalent to x(0). If a range tracking algorithm dictates that x(0) be x(399), that is, the range gate causes the PRI to be shortened, then y(397) and y(398) are calculated. If the range tracking algorithm dictates that x(0) be x(401), that is, the range gate causes the PRI to be increased, then x(399) and x(400) are calculated. The signal phase is preserved by using the correct x(0) and y(−2). The PRI is not limited to 4 μsec and can have a wide range of values. The filter algorithm is configured to set the N counter to count to 400 on the next cycle unless the range tracking algorithm requires 399 or 401 counts. In general, a filter configured similarly to filter  304  is capable of removing up to about 95% of the mathematical operations that are required in known filter processing schemes. 
     Another exemplary embodiment of filter  304 , for high altitude operation, incorporates a Barker code. Table 3 illustrates an exemplary embodiment, with a chip width equal to four, a PRI of 4 μsec, and P=400. In the exemplary embodiment, a 13 bit Barker code is used, and inputs x(0) and x(1) are data, x(2) and x(3) are filled with zeros, x(4) and x(5) are data, x(6) and x(7) are filled with zeros, and the pattern continues until N is equal to 51. Generally, the algorithm for N greater than 51 is given as y(N)=(−K 2 ) M  ×y(50), for N even, and y(N+1)=(−K 2 ) M ×y(51), where M=(N(even)−50)/2)−1. 
     
       
         
               
             
               
               
               
             
               
               
               
             
           
               
                 TABLE 3 
               
             
             
               
                   
               
               
                 Barker codes at high altitudes example 
               
             
          
           
               
                 x(N) 
                 Count (N) 
                 Algorithm 
               
               
                   
               
             
          
           
               
                 0 
                 397 
                 y(−3) = y(397) 
               
               
                 0 
                 398 
                 y(−2) = y(398) 
               
               
                 0 
                 399 
                 y(−1) = y(399) 
               
               
                 x(0) 
                 0 
                 y(0) = (1 / K1)[x(0) − x(−2)] − [K2 x y(−2)] 
               
               
                 x(1) 
                 1 
                 y(1) = (1 / K1)[x(1) − x(−1)] − [K2 x y(−1)] 
               
               
                 0 
                 2 
                 y(2) = (1 / K1)[x(2) − x(0)] − [K2 x y(0)] 
               
               
                 0 
                 3 
                 y(3) = (1 / K1)[x(3) − x(1)] − [K2 x y(1)] 
               
               
                 x(4) 
                 4 
                 y(4) = (1 / K1)[x(4) − x(2)] − [K2 x y(2)] 
               
               
                 x(5) 
                 5 
                 y(5) = (1 / K1)[x(5) − x(3)] − [K2 x y(3)] 
               
               
                 . 
                 . 
                 . 
               
               
                 . 
                 . 
                 . 
               
               
                 . 
                 . 
                 . 
               
               
                 0 
                 396 
                 y(−4) = y(396) = (−K2) 172  x y(50) 
               
               
                 0 
                 397 
                 y(−3) = y(397) = (−K2) 172  x y(51) 
               
               
                 0 
                 398 
                 y(−2) = y(398) = (−K2) 173  x y(50) 
               
               
                 0 
                 399 
                 y(−1) = y(399) = (−K2) 173  x y(51) 
               
               
                 x(0) 
                 0 
                 y(0) = (1 / K1)[x(0) − x(−2)]− [K2 x y(−2)] 
               
               
                 x(1) 
                 1 
                 y(1) = (1 / K1)[x(1) − x(−1)]− [K2 x y(−1)] 
               
               
                 0 
                 2 
                 y(2) = (1 / K1)[x(2) − x(0)]− [K2 x y(0)] 
               
               
                   
               
             
          
         
       
     
     FIG. 9 is a block diagram of a baseband IQ mixer  306 . Mixer  306  is configured to reject negative Doppler shifts on the IF (Intermediate Frequency) input signal, which are behind aircraft  2 , while allowing a positive doppler shift signal, from ahead of aircraft  2  to pass through. The positive doppler shift signal is equally forward as the negative doppler shift signal is behind. Referring specifically to mixer  306 , an IF in-phase portion includes a mixer  322  configured to operate at a frequency which is 1/PRI, where PRI is a radar pulse repetition interval, which converts the in-phase IF signal to Baseband (Doppler) frequency. Also included in the in-phase portion are a low pass filter  324 , a decimator  326 , and an all pass filter  328 . Referring specifically to mixer  306 , an IF quadrature portion includes a delay element  330 , which produces the IF quadrature signal, and a mixer  332  configured to operate at a frequency which is 1/PRI, where PRI is a radar pulse repetition interval, which converts the quadrature IF signal to Baseband (Doppler) frequency. Also included in the quadrature portion are a low pass filter  334 , a decimator  336 , and an all pass filter  338 . All pass filters  328  and  338  are configured to produce Baseband (Doppler) quadrature signals, which are received at a difference element  340 , where the output of the all-pass filter  338  is subtracted from the output of the all-pass filter  328 . The resulting difference signal contains the positive or forward-looking Baseband (Doppler) signal, which is received at swath bandpass filter  308 . 
     In particular embodiments, a frequency of data received at mixer  306  is 25 MHz, and is referred to as an IF (intermediate frequency) signal. Mixer  306  in one embodiment, is configured to convert the 25 MHz IF signal to baseband (or Doppler) frequencies, and further configured to reject negative Doppler frequencies. In specific embodiments, mixers  322  and  332  are configured with PRIs which allow decimation of the signal from correlation bandpass filter  304  to a 25 kHz sample rate. Specifically, in the embodiment shown, the allowed PRIs include 200, 400, 500, 800, and 1000. 
     For purposes of description, a current input to low pass filter  324  is given as x 1 (0). A current output of the low pass filter  324  is then given as y 1 (0)=(1/K 1 )[x 1 (0)+x 1 (−1)]−[K 2 ×y 1 (−1)], where x 1 (−1) and y 1 (−1) are respectively the previous input and output of the low pass filter  324 . A current input to low pass filter  334  is given as x 0 (0). A current output of the low pass filter  334  is then given as y 0 (0)=(1/K 1 )[x 0 (0)+x 0 (−1)]−[K 2 ×y 0 (−1)], where x 0 (−1) and y 0 (−1) are respectively the previous input and output of the low pass filter  334 . K 1  is 1+(1/tan(πfo/Fs 2 ), and K 2  is 1−(1/tan(πfo/Fs 2 ), where fo is bandwidth and Fs 2  is a sampling frequency of low pass filters  324  and  334 . In one embodiment, the sampling frequency of low pass filters  324  and  334  is the received signal frequency, Fs 1 , of 100 MHz divided by the pulse repetition interval. 
     The signals output from low pass filters  324  and  334  are further down sampled at decimators  326  and  336 . In one embodiment, decimators  326  and  336  are configured to sample at a frequency which is the pulse repetition interval multiplied by a sampling frequency, Fs 3 , of all pass filters  328  and  338 , divided by the received signal frequency, or (PRI×Fs 3 )/Fs 1 . 
     FIG. 10 is a block diagram  350  of Baseband (Doppler) in-phase all-pass filter  328  and Baseband (Doppler) quadrature all-pass filter  338 . In one embodiment, all-pass filter  328  and all-pass filter  338  include four cascaded second-order infinite impulse response (IIR) filters, configured to generate Baseband (Doppler) quadrature signals. Referring specifically to all-pass filter  328 , it includes filter elements  352 ,  354 ,  356 , and  358 , sometimes referred to herein as a, b, c, and d respectively. Referring to all-pass filter  338 , it includes filter elements  362 ,  364 ,  366 , and  368 , sometimes referred to herein as e, f, g, and h respectively. 
     FIG. 11 is a block diagram of one embodiment of a filter element  380 . Element  380  is a representation of all of filter elements  352 ,  354 ,  356 ,  358 ,  362 ,  364 ,  366 , and  368  (shown in FIG.  9 ). The following description refers specifically to element  380 , consisting of delay elements  392 ,  396 ,  400 ,  404 , summing element  386 , and gain elements  384 ,  394 ,  398 ,  388 ,  402 ,  406 . For the purposes of description the current input  382  is referred to as x(0). The current output  390  is then given as y(0)=[(A 0 *x(0))+(A 1 *x(−1))+(A 2 *x(−2))−(B 1 *y(−1))−(B 2 *y(−2))]/B 0 , where x(−1) and y(−1) are respectively the previous input and output of filter element  380 , and x(−2) and y(−2) are respectively the previous-previous input and output of filter element  380 . A 0 , A 1 , A 2 , B 1 , and B 2  refer to the gain block coefficients. 
     In one specific embodiment, the above equation is applicable for all of filter elements  352 ,  354 ,  356 ,  358 ,  362 ,  364 ,  366 , and  368  (shown in FIG.  9 ). The following are the coefficients for each filter element, the elements  352 ,  354 ,  356 ,  358 ,  362 ,  364 ,  366 , and  368  being represented by a, b, c, d, e, f, g, and h respectively, and BBfreq is the base band sampling frequency, and T is 1/BBfreq. In one embodiment, floating point precision is used. 
     Element a 
     a=1.0/0.3225; 
     w 0 =57.956; 
     A 2 =(4.0/T)/T+(2.0×w 0 ×a/T)+w 0 ×w 0 ; 
     A 1 =(−8.0/T)/T+2.0×w 0 ×w 0 ; 
     A 0 =(4.0/T)/T−(2.0×w 0 ×a/T)+w 0 ×w 0 ; 
     B 2 =(4.0/T)/T−(2.0×w 0 ×a/T)+w 0 ×w 0 ; 
     B 1 =(−8.0/T)/T+2.0×w 0 ×w 0 ; 
     B 0 =(4.0/T)/T+(2.0×w 0 ×a/T)+w 0 ×w 0 ; 
     Element b 
     b=1.0/0.4071; 
     w 0 =1198.2; 
     A 2 =(4.0/T)/T+(2.0×w 0 ×b/T)+w 0 ×w 0 ; 
     A 1 =(−8.0/T)/T+2.0×w 0 ×w 0 ; 
     A 0 =(4.0/T)/T−(2.0×w 0 ×b/T)+w 0 ×w 0 ; 
     B 2 =(4.0/T)/T−(2.0×w 0 ×b/T)+w 0 ×w 0 ; 
     B 1 =(−8.0/T)/T+2.0×w 0 ×w 0 ; 
     B 0 =(4.0/T)/T+(2.0×w 0 ×b/T)+w 0 ×w 0 ; 
     Element c 
     c=1.0/0.4073; 
     w 0 =16974.0; 
     A 2 =(4.0/T)/T+(2.0×w 0 ×c/T)+w 0 ×w 0 ; 
     A 1 =(−8.0/T)/T+2.0×w 0 ×w 0 ; 
     A 0 =(4.0/T)/T−(2.0×w 0 ×c/T)+w 0 ×w 0 ; 
     B 2 =(4.0/T)/T−(2.0×w 0 ×c/T)+w 0 ×w 0 ; 
     B=(−8.0/T)/T+2.0×w 0 ×w 0 ; 
     B 0 =(4.0/T)/T+(2.0×w 0 ×c/T)+w 0 ×w 0 ; 
     Element d 
     d=1.0/0.3908; 
     w 0 =259583.5; 
     A 2 =(4.0/T)/T+(2.0×w 0 ×d/T)+w 0 ×w 0 ; 
     A 1 =(−8.0/T)/T+2.0×w 0 ×w 0 ; 
     A 0 =(4.0/T)/T−(2.0×w 0 ×d/T)+w 0 ×w 0 ; 
     B 2 =(4.0/T)/T−(2.0×w 0 ×d/T)+w 0 ×w 0 ; 
     B 1 =(−8.0 T)/T+2.0×w 0 ×w 0 ; 
     B 0 =(4.0/T)/T+(2.0×w 0 ×d/T)+w 0 ×w 0 ; 
     Element e 
     e=1.0/0.3908; 
     w 0 =152.05; 
     A 2 =(4.0/T)/T+(2.0×w 0 ×e/T)+w 0 ×w 0 ; 
     A 1 =(−8.0/T)/T+2.0×w 0 ×w 0 ; 
     A 0 =(4.0/T)/T−(2.0×w 0 ×e/T)+w 0 ×w 0 ; 
     B 2 =(4.0/T)/T−(2.0×w 0 ×e/T)+w 0 ×w 0 ; 
     B 1 =(−8.0/T)/T+2.0×w 0 ×w 0 ; 
     B 0 =(4.0/T)/T+(2.0×w 0 ×e/T)+w 0 ×w 0 ; 
     Element f 
     f=1.0/0.4073; 
     w 0 =2326.03; 
     A 2 =(4.0/T)/T+(2.0×w 0 ×f/T)+w 0 ×w 0 ; 
     A 1 =(−8.0/T)/T+2.0×w 0 ×w 0 ; 
     A 0 =(4.0/T)/T−(2.0×w 0 ×f/T)+w 0 ×w 0 ; 
     B 2 =(4.0/T)/T−(2.0×w 0 ×f/T)+w 0 ×w 0 ; 
     B 1 =(−8.0/T)/T+2.0×w 0 ×w 0 ; 
     B 0 =(4.0/T)/T+(2.0×w 0 ×f/T)+w 0 ×w 0 ; 
     Element g 
     g=1.0/0.4071; 
     w 0 =32949.65; 
     A 2 =(4.0/T)/T+(2.0×w 0 ×g/T)+w 0 ×w 0 ; 
     A 1 =(−8.0/T)/T+2.0×w 0 ×w 0 ; 
     A 0 =(4.0/T)/T−(2.0×w 0 ×g/T)+w 0 ×w 0 ; 
     B 2 =(4.0/T)/T−(2.0×w 0 ×g/T)+w 0 ×w 0 ; 
     B 1 =(−8.0/T)/T+2.0×w 0 ×w 0 ; 
     B 0 =(4.0/T)/T+(2.0×w 0 ×g/T)+w 0 ×w 0 ; 
     Element h 
     h=1.0/0.3225; 
     w 0 =681178.9; 
     A 2 =(4.0/T)/T+(2.0×w 0 ×h/T)+w 0 ×w 0 ; 
     A 1 =(−8.0/T)/T+2.0×w 0 ×w 0 ; 
     A 0 =(4.0/T)/T−(2.0×w 0 ×h/T)+w 0 ×w 0 ; 
     B 2 =(4.0/T)/T−(2.0×w 0 ×h/T)+w 0 ×w 0 ; 
     B 1 =(−8.0/T)/T+2.0×w 0 ×w 0 ; 
     B 0 =(4.0/T)/T+(2.0×w 0 ×h/T)+w 0 ×w 0 ; 
     FIG. 12 is a block diagram of one embodiment of a swath band pass filter  308 . Filter  308  is a first order band pass filter which is centered on the doppler frequency. Filter  308  receives as input a signal, En, output from IQ mixer  306  (shown in FIG.  9 ). Further inputs include a filter center frequency in hertz, Fc, a filter bandwidth in hertz, B, and a filter sampling frequency in hertz, Fs, which are provided. 
     A filtered output signal, Eo, is determined according to Eo=(A 0 /B 0 )×En−(A 0 /B 0 )×En×Z −2 −(B 1 /B 0 )×Eo×Z −1 −(B 2 /B 0 )×Eo×Z −2 . Referring specifically to filter  308 , the input signal, En  422  is received and multiplied by a coefficient  424 , with a value of A 0 /B 0 , and then applied to a summing element  426 . The output of summing element  426  is filter output  428 . Input  422  is also delayed two counts by a two sample delay element  430  whose output is multiplied by coefficient  432 , with a value of −A 0 /B 0 , and then applied to summing element  432 . 
     Output  428  is multiplied by a sample delay element  434 , whose output is multiplied by a coefficient  436 , with a value of −B 1 /B 0 , and then applied to summing element  432 . Output  428  is also multiplied by a two sample delay element  438 , whose output is multiplied by a coefficient  444 , with a value of −B 2 /B 0 , and then applied to summing element  432 . Coefficients for filter  308  are determined according to Wb=2πB, which is bandwidth in radians, Wu=2π×(Fc+B/2), which is an upper 3 db point of filter  308  in radians, and W 1 =2π×(Fc−B/2), which is a lower 3 db point of filter  308  in radians. The coefficient A 0  is 2×Fs×Wb, B 0  is (4×Fs 2 )+(2×Fs×Wb)+(W 1 ×Wu), B 1  is (2×W 1 ×Wu)−(8×Fs 2 ), and B 2 =(4×Fs 2 )−(2×Fs×Wb)+(W 1 ×Wu). 
     FIG. 13 is a block diagram of a filter coefficients processor  309  (also shown in FIG. 7) which, in one embodiment, is configured to provide inputs to swath band pass filters  308  (shown in FIGS.  7  and  12 ). Processor  309  is configured to provide center frequencies Fc, for range swaths and phase swaths, and filter bandwidths, B, in hertz, for track and phase swaths and level and verify swaths. By controlling swath filter center frequencies, processor  309  is able to keep the doppler swath centered in the antenna beam. Also filter bandwidth is controlled. The filter bandwidth is directly related to a down track swath width on the ground such that a charge time for filter  308 , inversely but directly related to bandwidth, is equal to the time it takes aircraft  2  to fly across the swath width. Therefore, filter bandwidth is matched to velocity of aircraft  2 , and requires minimal processing. By knowing the antenna mounting angle, and the pitch of the aircraft, an angle to the antenna beam center is known, as described below, and a center frequency is calculated, generally, according to Fc=2×Velocity×sin(angle)/radar wavelength. 
     Referring specifically to processor  309 , an antenna mounting angle and velocity vectors in body coordinates are input to determine a doppler velocity, Vr  460 , at a range swath center frequency according to Vr=Vv×Cos(90-r-a)=Vv×Sin(a+r), where Vv=(Vx 2 +Vz 2 ) 0.5 , where Vx=velocity component on body x axis and Vz=velocity component on body z axis, a=A Tan(Vz/Vx), and r is the antenna mounting angle. A range swath center frequency, Fr  462  is determined according to Fr=2×Vr/L, where L is a wavelength, and in one specific embodiment, is 0.2291 feet. A velocity component on body y axis, Vy, is not used to center swath in antenna beam as the component has a value of zero since the antenna is fixed to a y axis of the body. 
     Processor  309  is also configured to determine a phase swath doppler velocity, Vp  464 , which is delayed behind the range swath by a time equal to the range processing delay. Vp is calculated as Vp=Vv×Cos(90-(r-p)-a)=Vv×Sin(a+r−p), where Vv=(Vx 2 +Vz 2 ) 0.5 , where Vx=velocity component on body x axis and Vz=velocity component on body z axis, a=A Tan(Vz/Vx), r is the antenna mounting angle, and p=(T×Vx/H)×(180/π) in degrees, where T=1/πB and is a delay through range swath filter, T×Vx is vehicle movement on body X axis, B is the swath bandwidth, and H is altitude in feet. Phase swath center frequency  466  is calculated according to Fp=2×Vp/L, where L is a wavelength, and in one specific embodiment, is 0.2291 feet. 
     Processor  309  is configured to determine a track and phase swath bandwidth, B  468  according to B=Vx/(0.6(H) 0.5 ) in hertz, where H is altitude in feet. A level and verify swath bandwidth  470  is calculated as a ratio of level and verify bandwidths to track and phase bandwidths, K, multiplied by track and phase swath bandwidth  468 . FIG. 14 is a vector diagram  500  which illustrates the calculations above described. In one embodiment, if the radar is in a range search mode, search range instead of altitude is used to calculate bandwidth. 
     Together, filters  308  and processor  309  automatically configure the radar doppler filter center frequency and bandwidth to achieve better radar performance over varying terrain and varying aircraft altitude, roll, and pitch than known systems. The determined center frequency operates to maintain the radar swath at an approximate center of the antenna beam. The calculated bandwidth is a bandwidth that controls the track swath width on the ground, and is calculated such that the filter time constant is equal to the time it takes the vehicle to move a corresponding swath width distance. The bandwidth corresponds to a time over the target and provides information as to how long a second swath lags a first swath. Phase channel swaths are set behind in position to account for a processing time of range processor  242  (shown in FIG.  7 ). The calculations of center frequency and bandwidth provide a mechanism for keeping a swath slightly in front of the aircraft such that a positive doppler shift is realized. 
     FIG. 15 is a block diagram of a phase processor  230  (also shown in FIGS.  6  and  7 ). Phase processor  230  includes three phase detectors  510 ,  512 , and  514 . In one embodiment, phase detectors  510 ,  512 , and  514  are configured with an input and a reference input, and further configured to determine a phase difference between the input and the reference input. Phase processor  230  is configured to receive processed radar return data, from swath band pass filters  308  (shown in FIG.  7 ), as described above, for all of a left channel, a right channel, and an ambiguous channel. Determination of phase difference in return data for the three channels allows for an accurate position determination for an object from which radar data was returned. 
     In the embodiment shown, phase detector  510  is configured to receive ambiguous channel return data as input, with left channel return data as a reference, and further configured to determine and output a phase difference between the left and ambiguous channels. Phase detector  512  is configured to receive right channel return data as input, with ambiguous channel return data as a reference, and further configured to determine and output a phase difference between the ambiguous and right channels. Phase detector  514  is configured to receive right channel return data as input, with left channel return data as a reference, and further configured to determine and output a phase difference between the left and right channels. 
     FIG. 16 is a block diagram of phase detector  510  (shown in FIG.  15 ). Phase detectors  512  and  544  are of the same configuration. Phase detector  510  incorporates a plurality of in-phase all pass filters  328  and quadrature all pass filters  338  (shown above in FIGS.  9  and  10 ). Specifically, an input is received at a first in-phase filter  520  (AP 1 . 1 ) and a first quadrature filter  522  (AP 1 . 2 ). A reference inputis received at a second in-phase filter  524  (AP 2 . 1 ) and a second quadrature filter  526  (AP 2 . 2 ). A multiplier  532  is configured to multiply outputs from filters  520  and  526 . Another multiplier  534  is configured to multiply outputs from filters  522  and  524 . A third multiplier  536  is configured to multiply outputs from filters  520  and  524 . A fourth multiplier  538  is configured to multiply outputs from filters  522  and  526 . An output of multiplier  534  is subtracted from an output of multiplier  532  with a subtraction element  540  which produces a Y output  542 . An output of multiplier  536  is added to an output of multiplier  538  with an addition element  544  which produces an X output  546 . A processing element  548  is configured to determine an arctangent of Y output  542  divided by X output  546 , which is the phase difference, in radians, between the input and the reference input. 
     In mathematical form, Y output  542  is calculated as Y=(AP 1 . 1 ×AP 2 . 2 )−(AP 1 . 2 ×AP 2 . 1 ), X output  546  is calculated as X=(AP 1 . 1 ×AP 2 . 1 )+(AP 1 . 2 ×AP 2 . 2 ), and the phase difference is A TAN (Y/X). 
     In one embodiment, in-phase filters  520  and  524  and quadrature filters  522  and  526  include the four cascaded second order infinite impulse response (IIR) filters as described in FIG.  10 . Further, in the embodiment, filters  520  and  524  are configured to include in-phase filter elements  352 ,  354 ,  356 , and  358 , (shown in FIG. 10) and are configured with coefficients which correspond to elements a, b, c, and d respectively as described above. Referring to quadrature filters  522  and  526 , they are configured to include quadrature filter elements  362 ,  364 ,  366 , and  368 , (shown in FIG. 10) and are configured with coefficients which correspond to elements e, f, g, and h respectively as described above. 
     Once phase differences between the right, left, and ambiguous channels has been determined, as described above, the phase differences are used, in one embodiment, to determine and interferometric angle to the target. FIG. 17 is a block diagram of phase ambiguity processing unit  232  (also shown in FIG.  6 ). In one embodiment, phase ambiguity processing unit  232  is configured to receive an electrical phase difference between the ambiguous channel and the left radar channel from phase detector  510 , an electrical phase difference between the right channel and the ambiguous radar channel from phase detector  512 , and an electrical phase difference between the right channel and the left radar channel from phase detector  514 . 
     Phase ambiguity processing unit  232  includes a phase bias adjust unit  570  which provides a phase shift value which compensates for phase shifts which occur in the routing of the radar signals, from receipt at an antenna and through cabling and processing areas within aircraft  2 . It is accepted that most phase shifting of signals occurs due to cabling for the routing of signals. Phase bias adjust  570  compensates for the ambiguous channel with respect to the left radar channel. Phase bias adjust  572  compensates for the right channel with respect to the ambiguous radar channel. Phase bias adjust  574  compensates for the right channel with respect to the left radar channel. 
     The compensated phase difference signals are received at a phase ambiguity resolver  576 . In one embodiment, phase ambiguity resolver  576  is implemented using software, and determines a physical (interferometric) angle to a target which originally reflected the radar signals received. Phase ambiguity resolution is further described below. After resolution of phase ambiguous signals, the physical angle signal is filtered utilizing a low-pass filter  578 , and an angular position of the target with respect to aircraft body coordinates (X,Y,Z) is determined from the physical angle to the target using body coordinates processor  233  (further described below). The determined position, in one embodiment, is 90 degrees minus a half angle of a cone whose axis is a Y-axis of the body of aircraft  2 . The target is on the cone surface, therefore providing the subtraction from 90 degrees above described. 
     
       
         
               
             
               
               
               
               
               
               
             
           
               
                 TABLE 4 
               
               
                   
               
               
                 Phase Ambiguity Resolution Matrix 
               
               
                   
               
             
             
               
                   
               
             
          
           
               
                 θ LA   
                 θ 1  = θ LA   
                 θ 1  = (θ LA  − 360) 
                 θ 1  = (θ LA  + 360) 
                   
                   
               
               
                   
                 Φ = sin −1 (θ 1 /K1 
                 Φ = sin −1 (θ 1 /K1 
                 Φ = sin −1 (θ 1 /K1 
               
               
                 θ AR   
                 θ 1  = θ AR   
                 θ 1  = (θ AR  − 720) 
                 θ 1  = (θ AR  − 360) 
                 θ 1  = (θ AR  + 360) 
                 θ 1  = (θ AR  + 360) 
               
               
                   
                 Φ = sin −1 (θ 1 /K2 
                 Φ = sin −1 (θ 1 /K2 
                 Φ = sin −1 (θ 1 /K2 
                 Φ = sin −1 (θ 1 /K2 
                 Φ = sin −1 (θ 1 /K2 
               
               
                 θ LR   
                 θ 1  = θ LR   
                 θ 1  = (θ LR  − 720) 
                 θ 1  = (θ LR  − 360) 
                 θ 1  = (θ LR  + 360) 
                 θ 1  = (θ LR  + 360) 
               
               
                   
                 Φ = sin −1 (θ 1 /K3 
                 Φ = sin −1 (θ 1 /K3 
                 Φ = sin −1 (θ 1 /K3 
                 Φ = sin −1 (θ 1 /K3 
                 Φ = sin −1 (θ 1 /K3 
               
               
                 θ LR   
                   
                   
                   
                 θ 1  = (θ LR  − 1080) 
                 θ 1  = (θ LR  + 1080) 
               
               
                   
                   
                   
                   
                 Φ = sin −1 (θ 1 /K3 
                 Φ = sin −1 (θ 1 /K3 
               
               
                   
               
             
          
         
       
     
     Table 4 is a phase ambiguity resolution matrix which is utilized, in one embodiment, to determine a physical angle to a target based upon electrical phase differences. A calculated electrical angle phase difference, θ, is equivalent to [(360×S)/λ]×sin(Φ) or K×sin(Φ), where Φ is the physical angle of the target in aircraft coordinates, S is a separation between the two antenna elements in feet, and λ is a wavelength of the radar signal in feet. In one particular embodiment, separation between the left antenna and the ambiguous antenna is 0.2917 feet (3.5 inches), separation between the ambiguous antenna and the right antenna is 0.7083 feet (8.5 inches), and the separation between the left antenna and the right antenna is 1 foot (12 inches). In the embodiment, the wavelength of the radar is 0.2291 feet. Therefore, in the embodiment, and referring to Table 4, K 1  is (360×0.2917)/0.2291, or about 458.4, K 2  is (360×0.7083)/0.2291, or about 1113.25, and K 2  is (360×1)/0.2291, or about 1571.64. Physical angles are then determined according to Φ=sin −1 (θ/K). 
     As antenna separation, radar wavelength, and aircraft position may all affect a timing of radar signals received at the various antennas, phase differences, which are determined as described above, will change at varying rates. In the embodiment illustrated in Table 4, physical angles are calculated for multiple electrical phase differences, and the true physical angle is a solution which provides approximately the same physical angle calculation, in each of the three rows (within a couple of degrees). Using the first antenna pairing (left and ambiguous), and based on antenna separation, three possible physical angles are determined from the electrical phase difference received from phase detector  510 . As the second antenna pairing (ambiguous and right) are further apart, five possible physical angles are determined. The last antenna pairing (left and right) are the furthest apart, therefore seven possible physical angles are determined. As described above, one of the physical angles from each group of physical angle calculations, will be roughly equivalent, thereby providing an unambiguous physical angle solution. In such a system it is important to note that separation in antenna pairing cannot be a multiple of radar wavelength. 
     FIG. 18 is a chart  600  illustrating varying electrical phase differences between three antenna pairings. Chart  600  helps to illustrate the process above described. As varying electrical phase differences between the three antenna pairings are charted, a single mechanical (physical) angle can be determined from the varying electrical phase difference plots for each antenna pairing. That is, for a physical angle, there is one solution which provides a phase difference for each radar channel grouping which is approximately equivalent to the calculated phase differences for the channel groupings. 
     FIG. 19 is a block diagram which illustrates inputs to and outputs from body coordinate processor  233  (also shown in FIG.  6 ). Processor receives the phase detector angle to the target from phase ambiguity resolver  576  via low pass filter  578  (described above in FIG.  17 ). Processor  233  further receives the doppler swath filter center frequency, and the filter bandwidth, a range to the target in feet, and velocity in pitch, roll and azimuth. Utilizing the processing described below, processor  233  is configured to determine a distance to the target in aircraft body coordinates. In one embodiment, the distance is determined in feet for aircraft body coordinates x, y, and z. Processor  233  further determines a velocity with respect to aircraft body coordinates in x and z. 
     FIG. 20 is a detailed block diagram of body coordinate processor  233  of FIG.  19 . Target range, vehicle velocity in pitch, roll, and azimuth, plus the swath filter center frequency and bandwidth are input into a doppler circle equation processor  620 , which is configured to determine doppler circle equations. The circle is determined using the swath filter center frequency equation Fc=[2×V×cos(β)]/L, where V is velocity, L is wavelength, and β is an angle with respect to a line of flight, which is determined through manipulation of the above equation. Therefore, β=cos −1 ((Fc×L)/(2×V)). A radius of the doppler circle, Rd, is calculated according to Rd=target range×sin(β). A distance of the doppler circle, Xd, from the aircraft is determined according to Xd=target range×cos(β). FIG. 21 is provided to illustrate the equations with regard to the doppler circle as derived above. 
     An example calculation is used to further illustrate. Inputs to doppler circle equation processor  620  include a range to target of 2000 feet, a velocity of 800 feet/second, a wavelength of 0.229 feet, and a doppler swath filter center frequency of 1213 Hertz. The angle with respect to the aircraft line of flight, β, is determined as b=cos −1 ((1213×0.229)/(2×800))=80 degrees. The doppler circle radius, Rd, is 2000×sin(80)=1969 feet, and distance of the doppler circle, Xd, is 2000×cos(80)=347 feet. 
     Again referring to FIG. 20, processor  233  further includes an interferometric circle equation processor  622  which is configured to determine interferometric circle equations in body coordinates. Processor  622  receives as input a target range and the interferometric angle (or phase detector angle), Φ, to the target as calculated by phase ambiguity resolver  576  (shown in FIG.  17 ). An interferometric circle radius, Ri, is calculated as Ri=target range×cos(Φ). A location of the interferometric circle on a Ym axis is determined as Ym=target range×sin(Φ). Referring to the example above, and including an interferometric angle input of 15 degrees, the radius of the interferometric circle, Ri, is 2000×cos(15), or 1932 feet. The location of the circle on the Ym axis, Ym is 2000×sin(15), or 518 feet. FIG. 22 is provided to illustrate the equations with regard to the interferometric circle as derived above. 
     Again referring to FIG. 20, a doppler to body coordinate transformation processor  624  within processor  233  uses the doppler circle equation, and pitch, roll, and yaw inputs to transform the doppler circle into body coordinates. Finally, at intersection processor  626  which is configured to solve equations to determine an intersection of the interferometric circle equation with the doppler circle equation that has been transformed into body coordinates. 
     In one embodiment, transforming begins by a determination of a velocity vector in body coordinates, from navigation data, N, (in pitch, roll, and yaw) according to                         V   X   N               V   Y   N               V   Z   N                          TRANSPOSE                 MATRIX            =                V   X   BODY               V   Y   BODY               V   Z   BODY                  ,                          
     where the transpose matrix is given by                     cos                   (   ψ   )                   cos                   (   θ   )               -   sin                     (   ψ   )                   cos                   (   θ   )             sin                   (   θ   )                   cos                   (   ψ   )                   sin                   (   θ   )                   sin                   (   φ   )       -     sin                   (   ψ   )                   cos                   (   φ   )                   -   sin                     (   ψ   )                   sin                   (   θ   )                   sin                   (   φ   )       -     cos                   (   ψ   )                   cos                   (   φ   )                 -   cos                     (   θ   )                   sin                   (   φ   )                   cos                   (   ψ   )                   sin                   (   θ   )                   sin                   (   φ   )       +     sin                   (   ψ   )                   cos                   (   φ   )                 cos                   (   ψ   )                   sin                   (   φ   )       -     sin                   (   ψ   )                   sin                   (   θ   )                   sin                   (   φ   )                 -   cos                     (   θ   )                   cos                   (   φ   )                  ,                          
     and ψ is azimuth, θ is pitch and φ is roll. 
     Velocity unit vectors (direction cosines) are given in body coordinates as Φ x =V x /(V x   2 +V y   2 +V z   2 ) 1/2 , Φ y =V y /(V x   2 +V y   2 +V z   2 ) 1/2 , and Φ z =V z /(V x   2 +V y   2 +V z   2 ) 1/2 . 
     Intersection processor  626  is configured to determine body coordinates which are calculated as X 1 =D×Φ x , Y 1 =D×Φ y , Z 1 =D×Φ z , where the velocity vector D, is given as R×cos(β), and β=cos −1 (Fc×L/2×V). B is the doppler cone angle, Fc is the swath filter center frequency, R is the range to the target, V is (V x   2 +V y   2 +V z   2 ) 1/2 , and L is the wavelength of the radar. 
     A position of the target in body coordinates is also calculated by intersection processor  626  as y=R×sin(A), where A=measured phase angle in body coordinates. The coordinate z is calculated as z=(−b±(b 2 −4ac) )/(2×a), where a=1+(Z 1 /K 1 ) 2 , b=(−4Z 1 ×KT/(2X 1 ) 2 ), and c=(KT/2X 1 ) 2 −KA. KA is calculated as (R×cos(A)) 2 , KB is calculated as (R×sin(B)) 2 , KY=(y−Y 1 ) 2 , and KT is calculated as KT=KA+KY−KB+X 1   2 +Z 1   2 . The coordinate x is calculated according to x=(KA− z   2 ) 1/2 . 
     In the above described systems, determination of the interferometric angle, Φ, is accomplished without regard to terrain radar return fading that is due to summations of out of phase radar returns from targets that are an extended distance from radar antennas  202 ,  208 , and  214  (shown in FIG.  6 ). FIG. 23 illustrates a unit  630  which applies filtering to the determined interferometric angle, Φ, to a target so that radar returns which exhibit such fading are weighted with respect to normal radar returns. Unit  630  includes phase processor  230  and phase ambiguity processor  232  as described above. Unit further includes a plurality of signal fade detectors  632  and a fade filter  634 . 
     Each signal fade detector  632  receives radar return data from one of the right, left, and ambiguous radar channels. Signal fade detectors  632  calculate a ratio of a signal envelope to an average signal for the radar channel to which it is connected. An output  636  of signal fade detectors  632  is a measure of depth of signal fade for the radar channel and provides a quality factor Q right , Q left , and Q ambiguous , respectively, for evaluation of the interferometric angle, Φ, as calculated by phase ambiguity processor  232  by fade filter  634 . 
     Therefore, signal fade detectors  632  and fade filter  634  provide filtering to the interferometric angle, Φ, resulting in adjustment of the effect of terrain features contributing to the interferometric angle, Φ, that is proportional (weighted) to a degree of radar return fading in the radar return signals resulting from the terrain features of the radar targets. Unit  630  then provides a corrected interferometric angle, Φ out , based at least in part on signal fade detectors  632  and fade filter  634 . 
     Referring to FIG. 23, the ratio of signal envelope for the radar returns to an average signal for the radar returns, Q fade , is calculated according to Q fade (n)=Q left (n)+Q right (n)+Q ambiguous (n), as the radar return includes a radar return for a right radar channel, a radar return for a left radar channel, and a radar return for an ambiguous radar channel, and “n” represents the nth pulse return. 
     Fade filter  634  then calculates a weighting for a received preliminary interferometric angle, Φ in (n), from phase ambiguity processor  232  according to              Φ   out          (   0   )       =       ∑     n   =     -   i         +   i                           Φ   in          (   n   )       ×         Q   fade          (   n   )       ÷       ∑     n   =     -   i         +   i                         Q   fade          (   n   )                 ,                          
     where i indicates a number of returns to be weighted in the filter. Assuming for example, a 25 point averaging filter a received preliminary interferometric angle, Φ in (n) is calculated according to:              Φ   out          (   0   )       =       ∑     n   =     -   25         +   25                           Φ   in          (   n   )       ×         Q   fade          (   n   )       ÷       ∑     n   =     -   25         +   25              Q   fade          (   n   )                 ,                where                     Φ   out          (   0   )                                
     is a filtered interferometric angle to a target, which is weighted based on an amount of radar return fade in the 25 radar return signals, and which allows navigation of a vehicle to be controlled based upon a weighting applied by the filter. 
     FIG. 24 provides a detailed illustration of one embodiment of signal fade detectors  632 . Signal fade detector  632  includes a device  640  which receives a radar return signal, and outputs a signal representative of an absolute value of the radar return signal. Signal fade detectors  632  also include a first low pass filter  642  with a cutoff frequency of 2 Hz, for example, that provides an average signal for the absolute value of the radar returns, sometimes referred to as a historic value for average return amplitude, and a second low pass filter  644  with a cutoff frequency of 300 Hz, for example, that provides an amplitude value of present returns, sometimes referred to as a signal envelope for absolute value of the radar returns. 
     A division device  646  provides the ratio of the radar return signal envelope to the average of the radar return signal, which is then provided to fade filter  634  (shown in FIG. 23) as described above. In a particular embodiment, outputs of low pass filters  642  and  644 , as well as output of division device  646  are calculated at a 25 kilohertz rate. In one embodiment, outputs of signal fade detectors  632  are coincident in time with preliminary interferometric angle, Φ in . Therefore the ratio of signal envelope to signal average and weighted interferometric angle, Φ out , are calculated utilizing the same radar return data. Which allows accurate calculation of body coordinates as described above. 
     While determining a position of a radar target with respect to, for example, an aircraft body, as described in detail above is necessary, it is also necessary in certain application to determine a range to a target. As is well known, in high altitude radar operations, it is possible that multiple radar transmit pulses will be transmitted before a return pulse is received. This is sometimes referred to as the ambiguous radar range problem. FIG. 25 illustrates one solution to the problem, the solution being to modulate radar transmit pulses  650  with a phase code. Implementation of the code, which involves a phase shifting of individual pulses of radar transmit pulses  650 , allows a synchronization of transmit pulses  650  with return pulses  652  which are received by a radar. Synchronization of the phase encoded radar pulses with the returned pulses is sometimes referred to as correlation. 
     In one embodiment, correlation is accomplished by implementation of a encoded radar scheme, and by looking for deviations in the return pulses from a reference, or starting altitude. FIG. 26 is a block diagram illustrating inputs to and outputs from range verification processor  244  (also shown in FIGS.  6  and  7 ). In one embodiment, verification processor  244  is configured to step through encoded return signals and determine a main lobe of the return signal to determine a range to, for example, a target. 
     Verification processor  244  is configured to receive as inputs, a detected radar return, which has been gated and demodulated. Verification processor  244  also receives as input a present internal range to the target, and a command from the radar search logic to be in either of a search mode or an acquisition mode. Verification processor  244  is configured with a variable mainlobe threshold factor (described below) and a verification dwell time, which is the time processor  244  is allocated to determine if an amplitude of a return signal exceeds the threshold factor. A verify status output is set true of the amplitude of the radar return exceeds the threshold value, thereby signifying that the transmit radar pulses and return radar pulses are correlated. If not correlated, the verify status output is false, and processor  244  provides a corrected range position to range processor  242  (shown in FIG.  7 ). 
     FIG. 27 is a flowchart  670  illustrating one embodiment of an autocorrelation process performed by processor  244 . Referring to flowchart  670 , a verify gate is set  672  to an internal range, from one of track or search. It is then determined whether a radar return is acquired  674  from within a verify gate, the gate attempting to align the chips of transmitted and received codes. If no target is acquired  674 , then processor  244  is configured to return to reset the verify gate. If a target is acquired  674 , then an amplitude of the return is determined  676 . In addition, the threshold factor is set to, for example, four times the determined amplitude and a counter is set to zero. The verify gate is stepped  678  out one chip of the code, the counter is incremented, and a dwell time passes before an amplitude of a return is again read. If the amplitude read is determined  680  not to be above the threshold factor, the counter is checked  682 . If the counter is determined to be less than one less than the number of chips within the barker code, the verify gate is again stepped  678 , and the steps are repeated, until the threshold factor is exceeded or the counter is equal to one less than the number of chips within the code. In one exemplary embodiment, a thirteen bit code is used, therefore the counter has a maximum value of twelve. In one embodiment barker codes are used for encoding the radar signals. 
     If the threshold factor is not exceeded, the original acquisition is an acquisition on the main lobe of the return, and the transmit and return codes are aligned, and the internal range as determined by processor  244  is correct, resulting in a verification status being set  684  to verify. 
     If the threshold factor is exceeded, then the transmit and return codes have become aligned. If the internal range has been moved  686  more than two range gates, the process illustrated by flowchart  670  begins anew. If there is a less than two range gate movement  686 , the search logic of the radar is set  688  to not verify, and is moved by the value of the counter, in order to align the transmit and receive barker codes. The process illustrated by flowchart  670  again begins. The continuous processing of encoded radar transmit and return signals by processor, provides a favorable solution to the known radar range ambiguity problem by constantly stepping through the codes to ensure receipt of an unambiguous radar range return. 
     In one embodiment, the above described verification processing for radar range ambiguity is applied continuously during flight, not just during initial acquisition. In utilization of such a system, the verification processing is applied in order to resolve range ambiguity during acquisition, but the processing is continuously applied after acquisition, throughout the flight. The continuous processing is done in order to ensure that if the transmit and received pulses become misaligned (loose correlation) the misalignment will both detected and corrected. Loss of correlation could occur due to, for example, a range discontinuity due to severe aircraft rolls or a sudden change in terrain (i.e. flying over a cliff). 
     The verification processing is further illustrated through an example. In one embodiment, a phase code is used to resolve radar range ambiguities and particularly a 13 bit phase code provides 20×log( 13 ) or 22 dB of rejection to range sidelobes. However, if verification processor  244  should, for some reason, line itself on an ambiguous side lobe, even if the mainlobe is for example 22 dB higher in amplitude, verification processor  244  will stay aligned with the sidelobe as long as there is a greater than 22 dB sensitivity margin. As stated above, one such example is flying over a sharp and deep cliff where a maximum radar track rate is less than a rate at which the range changes over the cliff. However, in practice, and assuming an ambiguous range sidelobe is lined up, a transition to a decreased sensitivity margin will normally result in a less than sufficient margin to track the ambiguous range side lobe. Examples include flying over poor reflectivity ground or encountering a severe aircraft roll. The result is verification processor  244  realigning into a proper and unambiguous line up onto the main lobe. Thus an ambiguous radar range does, after some time, normally correct itself However, and especially with auto pilot systems, severe and dangerous aircraft altitude corrections will result during the time of this very undesirable ambiguous range condition. 
     The method illustrated in flowchart  670  resolves the above illustrated situation by continuously searching for the main lobe, while tracking what is believed to be the correct position, or lobe. If during the ambiguity processing, or verification background search, it is determined that an ambiguous range is being tracked, an immediate correction is made to get the radar onto the correct range (i.e. the main lobe). To detect if the radar is on an ambiguous range track, the 20Log N equation is utilized to continuously determine differences between the main lobe, and undesired side lobes. 
     The above described methods and systems describe a digital signal processing solution to known radar target position and range ambiguity problems. Use of digital signal processing techniques therefore enables a radar system to perform faster and more accurate airborne processing than known radar ambiguity solutions. While the invention has been described in terms of various specific embodiments, those skilled in the art will recognize that the invention can be practiced with modification within the spirit and scope of the claims.