Abstract:
Devices and methods capable of correcting for distortion are disclosed. For example, a method for compensating for non-idealities in a frequency-conversion circuit having a high-frequency input side separated from a baseband side by a passive mixer is presented. The method includes injecting a plurality of calibration signals in the baseband side to determine cross-talk between an In-phase (I) portion of the baseband side and a Quadrature (Q) portion of the baseband side to produce a first measurement Γ +  and a second measurement Γ − , synthesizing a crosstalk compensation filter g(ω) based on the first measurement Γ +  and the second measurement Γ − , and applying the crosstalk compensation filter g(ω) to an output of the frequency-conversion circuit.

Description:
INCORPORATION BY REFERENCE 
     This application claims the benefit of U.S. Provisional Application No. 61/883,280 entitled “Compensation Algorithm for Baseband Filter Calibration” filed on Sep. 27, 2013, the content of which is incorporated herein by reference in its entirety. 
    
    
     BACKGROUND 
     Passive mixers are a highly desirable option when forming a mixer in a receiver. One drawback to using a passive mixer, however, is that passive mixers allow the components on each side (input and output) to interact. For example, an output impedance of a low noise amplifier (LNA) driving a passive mixer will interact with the input impedance of any filter coupled to the output of the mixer. As a result of such interaction, an ideal/desired transfer function of the filter(s) will be distorted from the ideal. 
     SUMMARY 
     Various aspects and embodiments of the invention are described in further detail below. 
     In an embodiment, a method for compensating for non-idealities in a frequency-conversion circuit having a high-frequency input side separated from a baseband side by a passive mixer is disclosed. The method includes injecting a plurality of calibration signals in the baseband side to determine cross-talk between an In-phase (I) portion of the baseband side and a Quadrature (Q) portion of the baseband side to produce a first measurement Γ +  and a second measurement Γ − , synthesizing a crosstalk compensation filter g(ω) based on the first measurement Γ +  and the second measurement Γ − , and applying the crosstalk compensation filter g(ω) to an output of the frequency-conversion circuit. 
     In another embodiment, a self-calibrating receiving circuit includes an amplifier with an input configured to be coupled to a high-frequency signal source and produce an amplified signal, a passive mixer coupled to an output of the amplifier and configured to split the amplified signal into an In-phase (I) signal at an I-signal output and a Quadrature (Q) signal at a Q-signal output, a first filter coupled to the passive mixer so as to receive the I-signal, a second filter coupled to the passive mixer so as to receive the Q-signal, signal injection circuitry configured to inject a first calibration signal followed by a second calibration signal into to first filter and the second filter so as to produce a first measurement signal Γ +  followed by a second measurement signal Γ − , and a crosstalk compensation filter g(ω) configured to remove crosstalk and coupled to an output of each of the I-signal output and the Q-signal output, the crosstalk compensation filter g(ω) being based on the first measurement Γ +  and the second measurement Γ − . 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Various embodiments of this disclosure that are proposed as examples will be described in detail with reference to the following figures, wherein like numerals reference like elements, and wherein: 
         FIG. 1  is a block diagram of an example receiving circuit capable of producing a baseband signal from a high-frequency signal. 
         FIG. 2  depicts a circuit equivalent to the receiving circuit of  FIG. 1 . 
         FIG. 3  depicts a modified circuit equivalent to the receiving circuit of  FIG. 1  whereby equivalent input signals are injected into the baseband side of the passive mixer. 
         FIGS. 4-5  depict the receiving circuit of  FIG. 1  modified so as to allow calibration signals to be injected into the baseband side of the passive mixer. 
         FIG. 6  depicts an architecture of filters useful to correct for parasitic crosstalk and non-idealities. 
         FIG. 7  is a flowchart outlining a set of example operations for providing compensating for a receiving circuit having a passive mixer. 
     
    
    
     DETAILED DESCRIPTION OF EMBODIMENTS 
     The disclosed methods and systems below may be described generally, as well as in terms of specific examples and/or specific embodiments. For instances where references are made to detailed examples and/or embodiments, it is noted that any of the underlying principles described are not to be limited to a single embodiment, but may be expanded for use with any of the other methods and systems described herein as will be understood by one of ordinary skill in the art unless otherwise stated specifically. 
     One of the most significant disadvantages of a passive mixer is its inherit bilateral nature. That is, passive mixers allow for the coupling of devices attached to it. For this reason a load input impedance of a mixer can affect the operations of a low-noise amplifier (LNA) driving the mixer. This situation can lead to critical issue for a Receiver (RX) front-end design. That is, if the LNA output impedance has a reactive component, the Local Oscillator (LO) upper frequencies and the LO lower frequencies will show an asymmetric transfer function. This means that for a received Radio Frequency (RF) signal, a portion of it&#39;s In-phase (I) energy is down-converted to the Quadrature (Q) baseband and vice-versa. This effect is particularly evident in a Frequency Translatable Impedance (FTI) based front-end where the baseband transfer function shows an amplitude droop that has to be mandatorily fixed or else signal down-conversion cannot be performed properly. Another possible issue is that the LNA output impedance can affect the baseband transfer function shape. 
       FIG. 1  is a block diagram of an example receiving circuit  100  capable of producing a baseband signal from a high-frequency signal, such as a cellular telephone communication signal. As shown in  FIG. 1 , the receiving circuit  100  includes a low-noise amplifier (LNA)  110 , a first passive mixer  120 , an I-baseband filter  130 , a second passive mixer  122 , a Q-baseband filter  132 , a local oscillator (LO)  140  capable of producing a local oscillation signal cos(ω LO  t), where ω LO  is the local oscillation frequency, and a phase shift device  142  capable of shifting the local oscillation signal cos(ω LO  t) by −π/2 radians. 
     In operation, a high-frequency signal V IN  is received and amplified by the LNA  110  so as to provide an amplified high-frequency signal to both the first passive mixer  120  and the second passive mixer  122 . 
     The first passive mixer  120  receives the amplified high-frequency signal and the local oscillation signal cos(ω LO  t), and mixes the amplified high-frequency signal to produce an I-phase signal V I  according to the equation:
 
 V   I   =I (½)cos(2ω 0   t )+ I (½)+ Q (½)sin(2ω 0   t ),  Eq.(1)
 
     Similarly, the second passive mixer  122  receives the amplified high-frequency signal and the local oscillation signal cos(ω LO  t)(shifted by −ω/2 radians), and mixes the amplified high-frequency signal to produce an Q-phase signal V Q  according to the equation:
 
 V   Q   =I (½)sin(2ω 0   t )+ Q (½)+ Q (½)sin(2ω 0   t ),  Eq.(2)
 
     The I-baseband filter  130  and the Q-baseband filter  132  then remove the high-frequency components such that only the I(½) and Q(½) components remain. 
     However, because the output of the LNA  110  can contain reactive components, such as parasitic capacitance and inductance, the V I  and V Q  signals will be contaminated with crosstalk such that a filtered output signal from the I-baseband filter  130  will produce an erroneous signal:
 
 V   I-OUT   =I (½)+ errQ,   Eq(3)
 
where errQ represents crosstalk from the Q-channel, and a filtered output signal from the Q-baseband filter  132  will produce an erroneous signal:
 
 V   Q-OUT   =Q (½)+ errI,   Eq(4)
 
where errI represents crosstalk from the I-channel.
 
       FIG. 2  depicts a circuit model equivalent to the receiving circuit  100  of  FIG. 1 . As shown in  FIG. 2 , the outputs of LNA  110  contain a parasitic resistor R OUT  and a parasitic capacitor C OUT . Because the capacitor C OUT  has memory, it is capable of producing a parasitic coupling between filters  130  and  132 . The high-frequency input Y IN  of  FIG. 1  is represented by current source I IN,RF , and as will be demonstrated below the current source I IN,RF  can be alternatively modeled. 
       FIG. 3  depicts a modified circuit model equivalent to the receiving circuit of  FIG. 1  whereby the high-frequency input current source I IN,RF  is replaced by an I-side baseband current source I IN-I-BB  and a Q-side baseband current source I IN-Q-BB  (=j*I IN-I-BB ).  FIG. 3  demonstrates that, rather than inject a high-frequency signal (for example, in the gigahertz range) to test for parasitic crosstalk, it is possible to perform the same test using less expensive and more easily realized baseband signals (for example, in the megahertz range) injected in the baseband side of the passive mixers  120  and  122 . 
       FIGS. 4-5  depict the receiving circuit  100  of  FIG. 1  modified so as to allow calibration signals to be injected into the baseband side of the passive mixers  120  and  122 . The local oscillator  140  and phase shift device  142  are removed from  FIGS. 4-5  for clarity. As shown in  FIG. 4 , two analog-to-digital (ADC) converters  430  and  432  are appended to I-baseband filter  130  and the Q-baseband filter  132 , respectively, with the output of the ADCs  430  and  432  provided to a processor circuit  450 . Additionally, two digital-to-analog circuits (DACs)  440  and  442  are provided as signal injection circuitry to inject baseband signals from the processor circuit  450  into the I-baseband filter  130  and the Q-baseband filter  132 , respectively. 
     In operation, DAC  440  injects a first calibration signal (cos(Δω t)) into the I-baseband filter  130  while DAC  442  injects a first complementary calibration signal (sin(Δω t)) into the Q-baseband filter  132 . The term Δω is equal to a desired test frequency, and injecting the cos(Δω t) signal into the I-baseband filter  130  and sin(Δω t) signal into the Q-baseband filter  132  is the equivalent of injecting a signal having a frequency f LO +f IN  into the input of the LNA  110 . The frequency Δω is a baseband frequency much less than frequency ω LO , where ω LO  is the local oscillator (LO) mixing frequency of the passive mixers  120  and  122 . As the first calibration signals are injected, the ADCs  430  and  432  produce a first measurement Γ +  to the processor circuit  450 . Generally, Γ(ω) is the output circuit response for any given input to the LNA  110 . For the purposes of this disclosure, the first measurement Γ +  is the response expected when a signal having a frequency f LO +f IN  is fed into the input of the LNA  110 . 
     Next, as shown in  FIG. 5 , DAC  440  injects a second calibration signal (cos(Δω t)) signal into the I-baseband filter  130  while DAC  442  injects a complementary second calibration signal (−sin(Δω t)) into the Q-baseband filter  132 , which is the equivalent of injecting a signal having a frequency f LO -f IN  into the input of the LNA  110 . As the second calibration signals are injected, the ADCs  430  and  432  produce a second measurement Γ −  to the processor circuit  450 . For the purposes of this disclosure, the second measurement Γ −  is the response expected when a signal having a frequency f LO -f IN  is fed into the input of the LNA  110 . 
     The processor circuit  450  then uses the first measurement Γ +  and the second measurement Γ −  to synthesize two different filters: a crosstalk compensation filter g(ω) and a transfer function correction filter h(ω). The correction filters are implemented by the correction filter circuitry  452  within the processor circuit  450 . The architecture of the filters can be found in  FIG. 6 , which shows two crosstalk compensation filters g(ω), two summing junctions and two transfer function correction filters h(ω). 
     The parameters of the crosstalk compensation filter g(ω) are determined according to the formula: 
                       g   ⁡     (   ω   )       =     j   ⁢         Γ   -   *     -     Γ   +           Γ   -   *     +     Γ   +             ,           Eq   .           ⁢     (   5   )                 
where j is the square-root of −1, and Γ* −  is the complex conjugate of Γ − .
 
     The crosstalk compensation filter g(ω) can be synthesized as any type of filter. However, as the order of the crosstalk compensation filter g(ω) is likely to be no greater than two, the crosstalk compensation filter g(ω) is well synthesized as any type of biquad filter, such as a Tow-Thomson biquad filter, and take the general form: 
                       g   ⁡     (   ω   )       =       jω   ⁢       Im   ⁡     [   Q   ]              Q        2             -     ω   2       +       jωω   0     ⁢       Re   ⁡     [   Q   ]              Q        2         +     ω   0   2           ,           Eq   .           ⁢     (   6   )                 
where j is the square-root of −1, Q is a quality-factor of Γ(ω), and ω 0  is the cut-off frequency of Γ(ω), assuming of Γ(ω) is a second-order low-pass function.
 
       FIG. 6  depicts the correction filter circuitry  452  of  FIGS. 4-5  useable to correct for parasitic crosstalk and non-idealities. As shown in  FIG. 6 , the correction filter circuitry  452  includes two crosstalk compensation filters g(ω), two summing junctions  610  and  612 , and two transfer function correction filters h(ω). In operation, the crosstalk compensation filters g(ω) and summing junctions  610  and  612  are applied to the I-portion and the Q-portion of the baseband side according to the formulae:
 
 I   BB-1   =I   BB   +g (ω) Q   BB   Eq. (7)
 
 Q   BB-1   =Q   BB   +g (ω) I   BB   Eq. (8)
 
where I BB  is an in-phase output of the passive mixer, Q BB  is a quadrature output of the passive mixer, I BB-1  is the in-phase output of the passive mixer compensated for crosstalk, and Q BB-1  is the quadrature output of the passive mixer compensated for crosstalk.
 
     The transfer function correction filters h(ω), which are also based on the first measurement Γ +  and the second measurement Γ − , are applied to the I-portion and the Q-portion of the baseband side according to the formulae:
 
 I   BB-2   =h (ω) I   BB-1   Eq(9)
 
 Q   BB-2   =h (ω) Q   BB-1   Eq(10)
 
where I BB-2  is the in-phase output of the passive mixer compensated for non-idealities, and Q BB-2  is the quadrature output of the passive mixer compensated for non-idealities.
 
     As with the crosstalk compensation filters g(ω), the transfer function correction filters h(ω), can be synthesized with any number of filters, such as a biquad filter. Equation (11) below shows a non-limiting example useful for the transfer function correction filter h(ω): 
                       h   ⁡     (   ω   )       =         -     ω   2       +     jω   ⁢       ω   0       Q   W         +     ω   0   2             -     ω   2       +     jω   ⁢       ω   0       Q   R         +     ω   0   2       ⁢                 ,           Eq   .           ⁢     (   11   )                 
where Q W  is the actual baseband quality factor taken from g(ω), and Q R  is a target filter quality factor.
 
       FIG. 7  is a flowchart outlining a set of example operations for compensating for non-idealities in a frequency-conversion circuit having a high-frequency input side separated from a baseband side by a passive mixer, such as that shown in  FIGS. 1-5  above. It is to be appreciated to those skilled in the art in light of this disclosure that, while the various functions of  FIG. 10  are shown according to a particular order for ease of explanation, that certain functions may be performed in different orders or in parallel. 
     At S 702 , where a first calibration signal (cos(Δω t)) is injected into an I-baseband filter while a complementary calibration signal (sin(Δω t)) is injected into a Q-baseband filter—both baseband filters being located on the baseband side of a passive mixer. As discussed above, this acts as the equivalent of injecting a signal having a frequency f LO +f IN  into the input of an amplifier on the high-frequency input side of the passive mixer. Next, at S 702  as the first calibration signals are injected, a first measurement signal Γ +  can be measured at the output of the baseband filters. The process continues to S 706 . 
     At S 706 , a second calibration signal (cos(Δω t)) is injected into the I-baseband filter while a complementary calibration signal (sin(Δω t)) is injected into the Q-baseband filter. As discussed above, this acts as the equivalent of injecting a signal having a frequency f LO -f IN  into the input of an amplifier on the high-frequency input side of the passive mixer. Next, at S 708  as the second calibration signals are injected, a second measurement signal Γ −  can be measured at the output of the baseband filters. The process continues to S 710 . 
     At S 710 , a crosstalk compensation filter g(ω) is determined and synthesized based upon Equations (5) and (6) above using the first measurement signal Γ +  and the second measurement signal Γ − . Next, at S 712 , a transfer function correction filter h(ω) is determined and synthesized based upon Equation (11) above using, for example, the first measurement signal Γ + , the second measurement signal Γ − , and quality factor of the crosstalk compensation filter g(ω). Then, at S 714  the crosstalk compensation filter g(ω) of S 710  and the transfer function correction filter h(ω) are applied as shown in  FIG. 6  and according to Equations (7) through (10) above. 
     While the invention has been described in conjunction with the specific embodiments thereof that are proposed as examples, it is evident that many alternatives, modifications, and variations will be apparent to those skilled in the art. Accordingly, embodiments of the invention as set forth herein are intended to be illustrative, not limiting. There are changes that may be made without departing from the scope of the invention.