Abstract:
An apparatus for providing Phased-Locked Loop (PLL) synthesis comprises a phase detector, at least one switchable filter, an oscillator controlled by a control voltage (u PLL ) and a divider. The controlled oscillator has two inputs, wherein the control voltage (u PLL ) is coupled to a first input, and a selection voltage (u SET ) for rough frequency adjustment is coupled to a second input. Both voltages establish the frequency of the oscillator.

Description:
FIELD OF THE INVENTION 
       [0001]    The present invention, according to various embodiments, relates to a PLL (phase-locked loop) synthesizer. 
       BACKGROUND OF THE INVENTION 
       [0002]    PLL (phase-locked loop) synthesizers are used in many contexts. For example, they are used in mobile-telephone technology, measurement technology and high-frequency measurement technology. PLL synthesizers are based on a reference oscillator or quartz oscillator, of which the frequency provides the basis for a specified output signal. A phase detector compares the reference frequency specified by the oscillator with the desired signal frequency and issues corresponding voltage pulses. These are then further processed in an integrator or filter circuit and supplied to a voltage-controlled oscillator (VCO). This adapts the frequency of the output oscillation according to the input specification, which, in the phase comparator or detector, is then supplied to the phase detector again via a feedback loop. 
         [0003]    On the basis of a high-precision reference frequency, any required frequencies can be generated by means of appropriate division factors. In this context, access is made to a phase and/or frequency control, wherein a version of the required phase function scaled by the factor n is compared with the oscillation of the oscillator. The goal of the control-loop design is to provide rapid frequency-change rates, low signal noise, low component costs and short transient times. 
         [0004]    In order to achieve rapid frequency selection in the context of DE 101 60 627 A1, an offset voltage is coupled via an additive-amplification circuit after a first filter stage. However, the disadvantage of this method is that with several selectable loop bandwidths, the cost of circuit technology is extremely high. 
       SUMMARY OF THE INVENTION 
       [0005]    Accordingly, there exists a need for providing a PLL synthesizer, wherein any possible additional noise input is prevented, and at the same time, the cost of the circuit is minimized. 
         [0006]    In accordance with one embodiment of the present invention, the features of claim  1  are provided. Further developments are specified in the dependent claims. 
         [0007]    In accordance with one embodiment of the present invention, further filter levels are avoided, and only passive circuit elements are used. Further filter levels for noise suppression have the additional disadvantage of having a direct influence on the loop dynamics, and, in the design of PLL synthesizers, they are also contrary to the optimisation of the loop dynamics and noise suppression. In an exemplary embodiment, a frequency-selection voltage is introduced as passively as possible and ideally directly in the VCO. This allows the above-named filter levels to be avoided, and a common treatment of the main noise-influencing factors and control dynamics is possible with only one filter. 
         [0008]    In accordance with one embodiment of the present invention realizes a circuit, which uses a VCO with only two inputs. Within the VCO, a voltage u SET  for rough frequency selection can be connected to a connection, and a control voltage u PLL  can be connected to a second input. The voltage u SET  for rough frequency selection can be previously filtered in this context. A control voltage u PLL , which is generated as previously by a phase detector by comparison of the input voltage and the frequency-divided output voltage, and of which the spectral properties are only determined through one filter, can be supplied to a second input. The associated VCO therefore has two inputs. Instead of adding the voltages u SET  and u PLL  to an active circuit and then filtering them again, the voltage for rough frequency selection is specified passively with this VCO realization directly in the VCO. 
         [0009]    In this manner, it is possible to obtain a control-circuit design simplified with regard to loop dynamics and noise behavior. With previous control-circuit structures, the filter design required the provision of two filters. With the PLL synthesizer according to the invention, the design is limited to one filter. There is therefore a complete absence of active components. The filter design is conceived in order to reduce the noise input of the reference oscillator and the phase detector and at the same time to provide an appropriate control dynamic. 
         [0010]    In accordance with one embodiment of the present invention, a noise-laden voltage for rough frequency selection u SET  can be filtered in an extreme manner without influencing the loop dynamics as a result. The required voltage u SET  is normally stored in a frequency-dependent manner in an electronic memory and is made available by a digital-analog converter. The switch is normally controlled in a digital manner, wherein the switching times are adapted to the desired loop bandwidth. The voltage u SET  specified in this manner is noise-laden and must be filtered. An R-C low-pass filter circuit pre-chargeable via a switch is used as the preferred circuit. As a result of a pre-charging of the capacitor of the low-pass filter circuit, the time constant can be bridged, and the filter can be used immediately for filtering the voltage u SET . On the one hand, the voltage is therefore immediately supplied in the full amplitude thereby providing an optimum switching voltage. 
         [0011]    On the other hand, higher-frequency noise components, which cannot be suppressed from the phase-control circuit, are blocked by a narrow-band filter. In this manner, rapid frequency-change rates can be achieved; the design of the loop dynamic can be made completely independent of the design of the filtering; and, at the same time, an extremely narrow-band filtering of the voltage u SET  can be achieved. 
         [0012]    Moreover, other methods for coupling a bias voltage are possible. A charging capacitor connected in parallel in a switchable manner to the capacitor of the R-C low-pass filter, which additionally smoothes the output voltage, can also be used. In this case also, other realizations can be used, for example, a second capacitor can be provided in parallel, in a switchable manner, with a time offset, so that the charging current does not exceed a maximum amplitude. 
         [0013]    Still other aspects, features, and advantages of the present invention are readily apparent from the following detailed description, simply by illustrating a number of particular embodiments and implementations, including the best mode contemplated for carrying out the present invention. The present invention is also capable of other and different embodiments, and its several details can be modified in various obvious respects, all without departing from the spirit and scope of the present invention. Accordingly, the drawings and description are to be regarded as illustrative in nature, and not as restrictive. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0014]    Various exemplary embodiments are illustrated by way of example, and not by way of limitation, in the figures of the accompanying drawings in which like reference numerals refer to similar elements and in which: 
           [0015]      FIG. 1  shows a schematic block-circuit diagram of a PLL synthesizer, according to an exemplary embodiment; 
           [0016]      FIG. 2  shows the schematic operation of a phase detector, according to an exemplary embodiment; 
           [0017]      FIG. 3  shows a block circuit diagram of a conventional PLL synthesizer with pre-tuning and selectable bandwidths conventional approach; 
           [0018]      FIG. 4  shows an exemplary embodiment of a PLL synthesizer, according to an exemplary embodiment; and 
           [0019]      FIG. 5  shows an exemplary embodiment of a VCO circuit, according to an exemplary embodiment. 
       
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       [0020]    Before describing the various embodiments of the present invention in greater detail, some basic principles of PLL synthesizers or respectively control circuits will first be explained below with reference to  FIGS. 1 and 2  as a basis for understanding the invention. An exemplary embodiment of the circuit according to the invention is then described with reference to  FIGS. 3 ,  4  and  5 . 
         [0021]    The principle of the phase-locked loop (referred to below as PLL) is based on a comparison of a reference signal with an actual signal or output signal. In this context, the reference signal is generated with a high-precision quartz oscillator. Using a programmable divider, it is possible to generate signals with a frequency, which can be specified within a wide framework. So-called phase detectors (referred to below as PD) or phase comparators are used for the comparison of the signals. 
         [0022]    The method of operation of a PD will first be modelled in a linear manner. This condition is fulfilled precisely when the loop is locked. Moreover, let us assume that the output voltage of the PD is proportional to the difference of the phases of the input signals, that is: 
         [0000]        u   d   =K   θ (θ i −θ o ) in [V]  (1) 
         [0000]    wherein θ i  and θ o  represent the phases of the reference-oscillator oscillation or the output signal. K θ  in this context is the proportionality factor of the phase detector and has the unit [V/rad] for phase detectors with voltage output or respectively [A/rad] for phase detectors with current output. The VCO is also assumed to be linear. The signal frequency at its output is obtained from the voltage at the filter output u e  [V] and from the gradient of the VCO characteristic K o  [Hz/V]. The output frequency Δω [rad/s] of the VCO can be specified as: 
         [0000]      Δ ω=K   o   u   e ( t )={dot over (θ)}( t )           K   o   U   e ( s )= s·Θ   o ( s )  (2) 
         [0023]    The VCO therefore has a low-pass filter character. A Laplace correspondence is indicated by          , and corresponding values in the time or respectively frequency domain are indicated by lower case or respectively upper case symbols for the corresponding characters in the formula, for example u e (t)         U e (s). The parameter s represents the frequency variable corresponding to the time variable t or respectively the dual frequency variable with unit [s −1 ]. Accordingly, the following equation is obtained for the transmission function B(s) of the closed control circuit: 
         [0000]    
       
         
           
             
               
                 
                   
                     B 
                      
                     
                       ( 
                       s 
                       ) 
                     
                   
                   = 
                   
                     
                       
                         
                           θ 
                           o 
                         
                          
                         
                           ( 
                           s 
                           ) 
                         
                       
                       
                         
                           θ 
                           i 
                         
                          
                         
                           ( 
                           s 
                           ) 
                         
                       
                     
                     = 
                     
                       
                         
                           K 
                           θ 
                         
                          
                         
                           K 
                           o 
                         
                          
                         
                           
                             F 
                              
                             
                               ( 
                               s 
                               ) 
                             
                           
                           / 
                           s 
                         
                       
                       
                         1 
                         + 
                         
                           
                             K 
                             θ 
                           
                            
                           
                             K 
                             o 
                           
                            
                           
                             
                               F 
                                
                               
                                 ( 
                                 s 
                                 ) 
                               
                             
                             / 
                             s 
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   3 
                   ) 
                 
               
             
           
         
       
     
         [0000]    wherein F(s) is the transmission function of a possible low-pass filter. In the simplest configuration, no low-pass filter is used, and B(s) is simplified to give: 
         [0000]    
       
         
           
             
               
                 
                   
                     B 
                      
                     
                       ( 
                       s 
                       ) 
                     
                   
                   = 
                   
                     
                       
                         
                           θ 
                           o 
                         
                          
                         
                           ( 
                           s 
                           ) 
                         
                       
                       
                         
                           θ 
                           i 
                         
                          
                         
                           ( 
                           s 
                           ) 
                         
                       
                     
                     = 
                     
                       
                         
                           
                             K 
                             θ 
                           
                           · 
                           
                             K 
                             o 
                           
                         
                         
                           s 
                           + 
                           
                             
                               K 
                               θ 
                             
                             · 
                             
                               K 
                               o 
                             
                           
                         
                       
                       = 
                       
                         1 
                         
                           1 
                           + 
                           
                             ( 
                             
                               s 
                               / 
                               W 
                             
                             ) 
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   4 
                   ) 
                 
               
             
           
         
       
     
         [0000]    which represents a low-pass filter characteristic with amplification 1 and bandwidth W=K θ ·K o . 
         [0024]    By contrast with the above model, a PD is based on the principle of converting input signals (generally sinusoidal oscillations) into rectangular signals by thresholding. A phase delay is then presented as a time delay of the edges of the rectangle. The difference between these mutually-displaced rectangular input signals gives a rectangular difference signal, which specifies the control voltage u PLL  as an adjustment value when converted into charge in the form of a pulse. The low-pass filter behavior for the VCO obtained in equation (2) integrates the voltage or charge pulses and gives the controlled phase θ o  of the output signal with the amplification factor K o . Within the VCO, the integration can be regarded as a derivation of a staircase function from the voltage pulses.  FIG. 1  additionally visualizes the difference formation of the input signals in the PD. 
         [0025]    A VCO includes an L-C oscillation circuit, in which, for example, in the given invention, capacitors are controlled in a voltage-based, adjustable manner. These controllable capacitors can be realised with variable-capacitance diodes or varactor diodes. Accordingly, the following applies for the known formula for the resonance frequency f R  of the oscillation circuit: 
         [0000]    
       
         
           
             
               
                 
                   
                     f 
                     R 
                   
                   = 
                   
                     
                       1 
                       
                         2 
                          
                         
                           π 
                           · 
                           
                             
                               L 
                               · 
                               
                                 C 
                                  
                                 
                                   ( 
                                   
                                     u 
                                     s 
                                   
                                   ) 
                                 
                               
                             
                           
                         
                       
                     
                      
                     
                         
                     
                      
                     
                       in 
                        
                       
                           
                       
                       [ 
                       Hz 
                       ] 
                     
                   
                 
               
               
                 
                   ( 
                   5 
                   ) 
                 
               
             
           
         
       
     
         [0000]    wherein C(.) is the characteristic of the voltage-controlled capacitor and u s  is the voltage for controlling the operating point. Accordingly, the following equation is initially obtained for the output frequency of the VCO: 
         [0000]    
       
         
           
             
               
                 
                   
                     
                       
                         
                           Δω 
                           R 
                         
                         = 
                           
                          
                         
                           2 
                            
                           π 
                            
                           
                               
                           
                            
                           
                             f 
                             R 
                           
                         
                       
                     
                   
                   
                     
                       
                         = 
                           
                          
                         
                           1 
                           
                             
                               L 
                               · 
                               
                                 C 
                                  
                                 
                                   ( 
                                   
                                     u 
                                     s 
                                   
                                   ) 
                                 
                               
                             
                           
                         
                       
                     
                   
                   
                     
                       
                         = 
                           
                          
                         
                           1 
                           
                             
                               L 
                               · 
                               
                                 C 
                                  
                                 
                                   ( 
                                   
                                     
                                       u 
                                       s 
                                     
                                     + 
                                     
                                       δ 
                                        
                                       
                                           
                                       
                                        
                                       u 
                                     
                                   
                                   ) 
                                 
                               
                             
                           
                         
                       
                     
                   
                   
                     
                       
                         ≈ 
                           
                          
                         
                           
                             K 
                             o 
                           
                            
                           δ 
                            
                           
                               
                           
                            
                           u 
                            
                           
                               
                           
                            
                           
                             in 
                              
                             
                                 
                             
                             [ 
                             
                               rad 
                               / 
                               s 
                             
                             ] 
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   6 
                   ) 
                 
               
             
           
         
       
     
         [0026]    With an appropriate linearization of (6) on the basis of the component characteristic C(.) about an operating point C(u s ), the VCO behavior can be modelled in a linear manner. 
         [0027]    The next section of this description describes the conventional approachconventional approach. As shown in  FIG. 3 , a PLL synthesizer A includes a PD (phase detector) B, to the inputs of which a reference frequency f ref =1/(2π){dot over (θ)} ref , for example, that of a reference oscillator, which is not shown, and the frequency f VCO /n=1/(2π){dot over (θ)} VCO /n, fed back via a programmable divider G via a feedback loop and divided down by the factor n, are connected. A first switchable filter C is provided at the output of the phase detector (PD) B. In the drawing, three possibilities are shown for the filter for broadband selection. Via an adding circuit D, a voltage for the rough frequency selection u SET  is added to the control voltage u PLL . The overall voltage is then filtered with a second filter E and supplied to the voltage-controlled oscillator F. As already indicated the output oscillation or respectively the output signal is then fed back via a divider G to the phase detector B. The output signal can be picked up at the output H. 
         [0028]    The control voltage u PLL  is connected at the output of the first filter C. At that position, an offset or switching voltage u SET  is added to the control voltage u PLL , so that a voltage u PLL +u SET  is obtained at the amplifier AMP. This resulting voltage is then amplified and filtered by means of a switchable filter E, in order to reduce the noise, which is superimposed over the voltage u PLL +u SET . The source of the noise is both from the voltage u SET  and also in the amplifier AMP, which acts as an adding unit. 
         [0029]    The transmission function of the phase detector PD is assumed to be two-dimensional and time-variant within the model. The transmission behavior of the PD can be modelled as linear at a certain point and for a certain duration, because, as can be seen in  FIG. 2 , the object can ultimately be understood as a modification of two sinusoidal input voltages by appropriate thresholding, which can be presented as a linear operation by multiplication with an appropriate function, in order to obtain by subtraction an appropriate output voltage, which communicates appropriate pulses to the integrator circuit F 1 (s), which converts the voltage pulses into mutually-displaced jump functions. 
         [0030]    In the realization according to the invention with an improved voltage-controlled oscillator (VCO), the voltage u SET  for rough frequency selection and the control voltage u PLL  are supplied directly to the VCO because of the advantages in circuit and control-technology. In this context, the object of the voltage u SET  is the rough frequency selection, that is to say, the rough pre-tuning to a required target frequency, and the object of control voltage u PLL  is to balance the remaining phase and/or frequency differences. 
         [0031]    The PLL synthesizer according to the invention, of which the exemplary embodiment is shown in  FIG. 4 , includes, as in the case of the conventional approachconventional approach, a phase detector  2 , a first filter  3  and a controlled oscillator (VCO)  4 , wherein a control voltage u PLL  is supplied to a first input, and a second voltage u SET  for frequency selection is supplied to a second input of the oscillator  4 . The filtering of the voltage u SET  for rough frequency selection takes place by means of a filter  6 , which is realised by an R-C low-pass filter, consisting of a resistor  6   1  and a capacitor  6   2 . A switching element  6   3 , which bridges the resistor  6   1 , allows a rapid pre-charging of the capacitor  6   2 . The capacitor  6   2  can advantageously be adjustable or can consist of two individual capacitors, wherein one of these individual capacitors can be added to the system by switching. 
         [0032]    By contrast with the conventional approach, the voltage u SET  for rough frequency selection is not supplied within the control circuit, but is supplied to the controlled oscillator (VCO)  4  at a second input. Accordingly, a second filter is not required, because the voltage u SET  for rough frequency selection, which originates from a digital/analog converter (not shown) and is filtered outside the control circuit with the low-pass filter  6 , is supplied directly to the oscillator (VCO)  4  at its second input. The inherent time constant of the filter  6  can be bridged by using the switching element  6   3  and therefore allows faster frequency-change rates by comparison with the conventional approachconventional approach. 
         [0033]      FIG. 5  shows an exemplary embodiment of the resonator of a voltage-controlled oscillator.  FIG. 5  shows only a part of a VCO, namely the resonator. A complete VCO is provided only if an active component (transistor) is added. The oscillator  4  includes two adjustable capacitors  4   2  and  4   4 , preferably diodes, especially varactor and/or variable-capacitance diodes, which form an oscillation circuit with a fixed inductance  4   6 ; an output capacitor  4   7 , which forms a short-circuit for high-frequency voltages and currents and blocks any constant and/or low-frequency current and/or voltage components; and a capacitor  4   5 , which forms a short-circuit to earth for high-frequency signals. 
         [0034]    In particular, a first resistor  4   1  is connected by a second connection to a first connection of a first adjustable capacitor, especially a variable-capacitance diode and/or varactor diode  4   2 , and a second resistor  4   3  is connected by a first connection to a second adjustable capacitor, in particular a variable-capacitance diode and/or varactor diode  4   4 . The second diode  4   4  in this context is connected by a second connection to a second connection of the resistor  4   1  and to a first connection of the first diode  4   2 . The second resistor  4   3  is connected via the capacitor  4   5  to earth. A first connection of the inductance  4   6  is connected to a second connection of the second resistor  4   3 , to a first connection of the capacitor  4   5 , to a first connection of the second diode  4   4  and to a second connection of the first diode  4   2 . 
         [0035]    The inductance forms a parallel circuit with the diodes  4   2  and  4   4 . The throughput direction of the second diode  4   4  extends in this context from a second connection of the first resistor  4   1  to a second connection of the second resistor  4   3 . The throughput direction of the first diode  4   2  extends from a second connection of the resistor  4   1  to a second connection of the inductance  4   6 . The inductance  4   6  and the first diode  4   2  are connected in each case to a second connection at the point  4   8  via a capacitor  4   7  to the output  4   9  of the resonator. A first connection of the inductance  4   6 , which is connected to a second connection of a second resistor  4   3  and to a first connection of the second diode  4   4 , can also be connected directly to earth. 
         [0036]    The main elements of the oscillation circuit are formed from the voltage-controlled diodes  4   2  and  4   4  and the inductance  4   6 , wherein the voltages u PLL  and respectively u SET  are coupled via a first connection of the resistors  4   1  and respectively  4   3 . The capacitor  4   5  in this context forms a direct-voltage separation to earth and a through connection for high-frequency oscillations. By contrast, the output capacitor  4   7  additionally decouples direct voltage and low-frequency oscillation or signal components towards the output  4   9  from the output voltage u A . 
         [0037]    The voltages u PLL  and u SET  adjust the variable-capacitance diodes  4   2  and  4   4  to an operating point dependent upon specifications from the phase detector and the frequency selection. In this context, the voltage difference Δu 1 =u PLL −u A ′ determines a first capacitance on the basis of the characteristic C v     1   (Δu 1 ) of the first diode  4   2 , and the voltage difference Δu 2 =u PLL −u SET  determines a second capacitance on the basis of the characteristic C v     2   (Δu 2 ) of the second diode  4   4 . The first voltage difference in this context is identical to the second voltage difference, because the inductance  4   6  for constant voltage u SET  represents a through connection, that is to say u A ′=u SET  and accordingly Δu 2 =Δu 1 . With this VCO circuit, a direct earth can also be provided, instead of the capacitor  4   5 . 
         [0038]    The total capacitance [C ges ] of the varactor diodes  4   2  and  4   4  in this context can be calculated after pre-tuning of the voltage using the following formula: 
         [0000]    
       
         
           
             
               
                 C 
                 ges 
               
                
               
                 ( 
                 
                   
                     Δ 
                      
                     
                         
                     
                      
                     
                       u 
                       1 
                     
                   
                   , 
                   
                     Δ 
                      
                     
                         
                     
                      
                     
                       u 
                       2 
                     
                   
                 
                 ) 
               
             
             = 
             
               
                 
                   
                     C 
                     
                       v 
                       1 
                     
                   
                    
                   
                     ( 
                     
                       Δ 
                        
                       
                           
                       
                        
                       
                         u 
                         1 
                       
                     
                     ) 
                   
                 
                  
                 
                   
                     C 
                     
                       v 
                       2 
                     
                   
                    
                   
                     ( 
                     
                       Δ 
                        
                       
                           
                       
                        
                       
                         u 
                         2 
                       
                     
                     ) 
                   
                 
               
               
                 
                   
                     C 
                     
                       v 
                       1 
                     
                   
                    
                   
                     ( 
                     
                       Δ 
                        
                       
                           
                       
                        
                       
                         u 
                         1 
                       
                     
                     ) 
                   
                 
                 + 
                 
                   
                     C 
                     
                       v 
                       2 
                     
                   
                    
                   
                     ( 
                     
                       Δ 
                        
                       
                           
                       
                        
                       
                         u 
                         2 
                       
                     
                     ) 
                   
                 
               
             
           
         
       
     
         [0000]    wherein Δu 1 =u PLL −u A ′ represents the voltage drop across the first diode  4   2  and Δ u2 =u PLL −u SET  represents the voltage drop across the second diode  4   4 . In equation (7), C v     1   (Δu 1 ) represents a characteristic of the first diode  4   2 , and C v     2   (Δu 2 ) represents a characteristic of the second diode  4   4 . The voltages u SET , u PLL  and u A ′ in this context are voltages for rough frequency selection (u SET ) and for fine tuning (u PLL ) and/or balancing of the operating point and a voltage (u A ′) at the point  4   8  of the VCO circuit. 
         [0039]    The following section of the description describes further advantages of the embodiment according to the invention shown in  FIG. 4 . 
         [0040]    The use of a charge pump (not shown), means that active filters can be completely dispensed with in the realization according to the invention. Noise inputs therefore only occur as a result of noise components or interference-frequency components in the oscillator reference signal and passive components. 
         [0041]    The filter  3  is designed as a stepped low-pass filter. A low-pass-filter behavior is achieved in cooperation with the properties of the remaining loop elements. Appropriate component values for an R-C circuit can be determined using an automated design method. 
         [0042]    The control voltage u PLL  may have a variance range, for example, of ±1.5 V, as a result of which, the form of the output oscillation changes to a minimal extent in accordance with the component characteristics. 
         [0043]    The present realization uses voltage-controlled oscillator circuits and varactor diodes. However, other known alternative oscillator circuits and frequency-determining elements can also be used.