Abstract:
A coupling device in FIG.  3  consisting of upper and lower connecting plates  100  and  101  with external flanges parallel to transmission line ( 103 ) for coupling RF energy for forward power detection. The coupling device ( 100 ) incorporates a helix structure with rotation centered near or about transmission line  103  and incorporates embedded secondary structures which are parallel to transmission line and fixed a predetermined distance from the transmission line ( 103 ). Theses plurality of parallel flanges are used to increase the coupling coefficient and directivity of the helix coupler ( 107 ) and maintain geometries that optimize magnetic field coupling. One or more vias ( 102 ) are used to connect individual upper connecting plate ( 100 ) and individual lower connecting plate ( 101 ) to form the overall helix structure. The addition of the parallel flanges to upper and lower connecting plates allow for a greater coupling efficiency per unit length of transmission line  103.

Description:
TECHNICAL FIELD 
     This invention relates in general to two-way radios and more particularly to RF power coupling. 
     BACKGROUND 
     Radio Frequency (RF) Coupling networks are well known in the art and are used in many different applications. These applications include transmit power control, radar detection and control, isolation and feedback strategies. Particular to the portable radio, coupler designs are critical to optimizing transmit output power while providing protection to the transmit power amplifier. A properly designed coupler will provide differentiation between incident and reflected RF energy (directivity) while exhibiting uniform coupling efficiency over the desired RF frequency range. The physical design of the coupler must accommodate well understood relationships between RF voltage potentials and their associated electric fields, and RF current densities and its associated magnetic fields. Both magnetic (H-field) and electric fields (E-field) are present in a propagating RF signal; however, the coupler&#39;s physical design can be constructed to primarily operate off either field (electric or magnetic) as the fundamental coupling mechanism. The relationship between the two fields are defined by a family equations known as Maxwell&#39;s equations. The electromagnetic vector relationships germane to this discussion are shown in Table 1 below: 
     
       
         
               
             
               
               
               
             
               
             
               
               
               
             
           
               
                 TABLE 1 
               
               
                   
               
             
             
               
                 S = E × H 
               
             
          
           
               
                 where 
                 S = 
                 Power per unit area (W/m{circumflex over ( )}2)-Poynting Vector 
               
               
                   
                 E = 
                 Electric Field Intensity Vector 
               
               
                   
                 H = 
                 Magnetic Field Intensity Vector 
               
             
          
           
               
                 and 
               
               
                 V × H = J + ∂D/∂t; 
               
             
          
           
               
                 where 
                 V = 
                 Direction of the axis of rotation and the magnitude of the 
               
               
                   
                   
                 rotation Vector 
               
               
                   
                 H = 
                 Magnetic Field Intensity Vector 
               
               
                   
                 J = 
                 Current Density Vector 
               
               
                   
                 D = 
                 Magnetic Flux Density 
               
               
                   
                 ∂/∂t = 
                 time derivative 
               
               
                   
               
             
          
         
       
     
     From Maxwell&#39;s equations, it is shown that a time varying electric and magnetic field generated by RF current densities or voltage potentials applied to a transmission line induce electromagnetic (EM) fields in the surrounding region. If a second conductor is positioned within the region, the magnetic field (H-field) will induce a current vector J proportional to the conductor&#39;s surface area which is perpendicular to propagating EM fields 
     The vector direction of the induced field is determined by the Poynting Vector S. This principle holds true for both co-planar and offset structures that are positioned on different planes from the main transmission line. By using the “Right Hand Rule” to S, the direction of the current flow within a given conductor can be determined by rotating the index finger of the right hand from the E-field vector to the H-field vector, and noting that the extended thumb points in the direction of the current flow. A diagram of a transmission line with a coupling structure positioned in an upper plane (not directly over) the main transmission line is shown in  FIG. 1 . 
     Prior art  FIG. 1  shows the top surface of the transmission line  101  exhibits a magnetic field polarity that is “additive” to the H-field of the lower surface Upper Plate  100  (coupling structure). For this condition to exists, the induced current “i o ” in the upper plate  100  must flow in the OPPOSITE direction of the current “i 1 ” in transmission line  101 . If the plate current were “forced” to flow in the same direction as the transmission line current, the H-fields between the transmission line and the plate would tend to “cancel” each other and coupling efficiency would be degraded. Thus, any geometry that enhances the coupler structure performance based on magnetic field coupling must provide complementary magnetic fields with the same polarity as that distributed on the transmission line. 
     It is also apparent from  FIG. 1  that E-fields are perpendicular to the H-field. Thus, for the E-fields to couple efficiently, the coupling structure should “overlap” the transmission line to maximize the conductor surface area which is perpendicular to the E-fields. This method of coupling can be described as Capacitive coupling (or broad edge coupling). This is different from a structure that utilizes magnetic field coupling, since maximizing perpendicular surface area to the H-field only requires a parallel structure to the transmission line, it does not necessitate overlap. These principles hold true for any three-dimensional (3-D) structure, including those which may be embedded into a PC board. 
       FIG. 2  illustrates a 3-D multi-plane coupling structure of the prior art with the associated upper and lower coupling plates  200  and  202  with current vectors (“i o ” and “i 2 ”, respectively). The coupling plates  200  and  202  have an H-field generated by RF current “i i ” propagating along transmission line  201  that induces H-fields of like polarity on coupling plates  200  and  202  for the surface facing the transmission. In other words, the H-field for the lower side of upper plate  200  is vectored the same as the top side of transmission line  201 , and the H field for the top side of lower plate  202  is vectored the same as the lower surface of transmission line  201 . These time varying magnetic fields generate current vectors i o  and i 2  respectively. It is important to note that the H-field polarity of the lower surface of upper plate  200  CONFLICTS with the top surface polarity of lower plate  202 . Thus, regions where the upper and lower plate overlap will cause reduced coupling efficiency, due to H-field cancellation, if they are electrically connected together (i.e. Current flows in the same direction in both structures). It is therefore imperative that the multi-layered helical geometry minimize regions of overlap between differing planes having current vectors oriented in the same direction. 
     From the foregoing discussion, it is evident that the two fields (E-field and H-field) are complementary, but independent mechanisms that can be used to provide coupling of RF energy. The electromagnetic relationship between them is determined by Maxwell&#39;s equations, which form guidelines that must be met to maximize coupling efficiency while providing superior directivity. 
     Directivity is a measure of the reflected and incident field differentiation of a coupler. The algebraic difference in decibels (dB) of the forward and reverse coupling coefficients for any fixed structure is defined as the directivity of that structure. A 20 dB directivity factor is considered acceptable for a bi-directional coupler. Historically, high performance bi-directional couplers have been fabricated on a substrate such as alumina, with thin film processes and their tight tolerance capability defining the coupler geometries to achieve a controlled 20 dB coupling coefficient with greater than 20 dB of directivity. Although the modularized approach to implementing the coupler is effective, it adds cost and process steps that could otherwise be eliminated if the coupler were to be embedded into the printed circuit board (PCB). To achieve high reliability coupler performance with existing PCB make tolerances of ±2 mil gaussian width distributions requires an innovative design approach. This innovation is based on selecting the proper coupling mode (E-field or H-field) that provides a design meeting manufacturing and performance requirements, while minimizing cost and area. Therefore, the need exists for embedded PCB coupler structure which achieves these desired objectives. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a profile view of magnetic &amp; electric field polarities for single plate coupler structure &amp; main transmission line used in the prior art. 
         FIG. 2  is a profile view of magnetic &amp; electric field polarities for multi-plane coupler structure &amp; main transmission line used in the prior art. 
         FIG. 3  is a 3-dimensional profile view of the preferred embodiment of the invention, illustrating a planar helix structure wrapped around the main transmission line with secondary coupling appendages parallel to the transmission line. 
         FIG. 4  is a top view of helix structure of  FIG. 3  (YAG Structure) with ports  1 ,  2 ,  3 , and  4  designating inter circuit connections. Three dimensional axis designation X, Y and Z are also indicated with the Z plane bisecting the page. 
         FIG. 5  is a top view of a helix structure with parallel structure within the Z plane at vias. (C Structure) with ports  1 ,  2 ,  3 , and  4  designating inter circuit connections. Three dimensional axis designation X, Y and Z are also indicated with the Z plane bisecting the page. 
         FIG. 6  is a top view of a second embodiment of the invention maximizing parallel coupling surface area within the XY plane embedded within a multi layered Helix structure. (BOW Structure) with ports  1 ,  2 ,  3 , and  4  designating inter circuit connections. Three dimensional axis designation X, Y and Z are also indicated with the Z plane bisecting the page. 
         FIG. 7  is a side view of the multi-layer structure showing dielectric thickness and plane designation of the helix/structure shown in  FIGS. 3 through 6  Note that dielectric thickness  701 ,  702 ,  703 , and  704  may vary independently depending on the particular coupling structure topology used and the performance target. 
         FIG. 8  is a 3-dimensional profile view of yet another embodiment of the invention, illustrating a continuous helix structure wrapped around the main transmission line with secondary coupling elements parallel to the transmission line. 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     The focus of this discussion is to detail a methodology of magnetic field coupling and the associated geometric constraints. A summary of the electrical constraints that maximize magnetic field coupling is listed below. 
     1) The maximally efficient inductive structure is a closely wound helical geometry where H-fields from contiguous windings are additive. This structure provides the smallest-(most spatially efficient) three-dimensional resonant structure and can be used to induce a band pass coupling response. 
     2) For multi-layer planar structures, minimize H-field cancellation by minimizing overlap between coupling structures that are also electrically connected to each other. 
     3) Since current densities are maximal at the EDGES of a planar transmission line, maximum H-field coupling and directivity performances achieved by maximizing the coupling structure&#39;s EDGE surface area that runs PARALLEL to the main transmission line. 
     4) Ensure that overall current flow induced onto the coupling structure is not “folded back” onto itself by utilizing structures that have 180° discontinuities within the same plane. 
     Some of these guidelines may tend to conflict with each other (i.e.: #1 &amp; #3 above); however, optimization for best performance is still realizable. This is achieved by adopting a helix as the basic geometric structure used in the coupler, but with sufficient space between contiguous windings to allow secondary structures to be embedded to maximize the magnetic (H-Field) coupling mode. Since planar structures are configured based on the frequency of operation, spatial requirements and desired performance for a given application, a final element of the embodiment includes series lumped element components as resonant elements. In summary, the overall concept as disclosed in this invention is:
         the implementation of unique multi-planar geometry(s) incorporating a helix topology “wrapped” around main transmission line (Helix with transmission line running down center axis), which is adaptable to being embedded into a RF PCB:   Coupling plates embedded into helix to maximize magnetic field (inductive) mode coupling; and   Series lumped element capacitor to achieve band pass resonant response; also serving to block direct current (DC) from the rectifying diode.       

     Referring now to the preferred embodiment shown in  FIG. 3 , a stripline transmission line  301  is positioned along the center axis of a multi-planar helical structure comprised of upper plate  300  and lower plate structures  302  electrically connected together by a series of vias  303 . Structures  300 ,  301 , and  302  are all on different planes, each plane separated by a dielectric material with characteristic impedance and velocity factor characteristics selected for the particular RF application.  FIG. 7  illustrates this planar relationship for stripline structures, with structures  300 ,  301 , and  302  of  FIG. 3  being located on Layer  1 , Layer  2  and Layer  3  of  FIG. 7  respectively. The ground planes (not shown) are positioned above and below the entire structure in compliance with classical stripline techniques. Note that passive elements equivalent to capacitor  304  and resistor  305  of  FIG. 3  are not shown in  FIG. 7 . For classical microstrip transmission line designs, a single ground plane (not shown) is positioned below lower plate structures  302  and air dielectric is used above the transmission line  301 ; thus the upper plate structures  300  are functionally “air bridges” over transmission line  301 . 
     Transmission line  301  is generally attached to an RF source such as an RF power amplifier (not shown) and provides for efficient propagation of RF energy within a specific frequency range to subsequent stages such as harmonic filters (also not shown). As RF power is transmitted down the main transmission line, EM near-fields are generated which are coupled to adjacent structures  300  and  302 . The physical dimension of the transmission line  301  is determined through classical methods dependent on substrate dielectric constants, desired characteristic impedance, and distance from ground planes (not shown). The physical dimensions of upper and lower coupling plates  300  and  302  are set to minimize capacitive (E-field) coupling; thus overlap with either the main transmission line or with subsequent interconnected plates are minimized. This means that the coupling plated are generally thin with respect to the transmission line (high impedance) and designed to maximize surface area parallel to the main transmission line (perpendicular to the H-field) through using secondary appendages or geometric structures. The number of turns of the helix structure is set by the coupling factor and frequency of operation required for the given application. The greater the number of turns for a given frequency, the higher the coupling factor. The higher the frequency of operation, the lower the number of turns required for a given coupling factor. 
     The direction of the power detected for the coupler is determined by the location of the terminating impedance (for example the resistor  305  in  FIG. 3 ). Given forward RF power is defined as propagating along the transmission line  301  from left-to-right as indicated in  FIG. 3 , then coupled RF energy proportional to the forward power will propagate through capacitor  304  when terminating resistor  305  is positioned as shown in  FIG. 3 . If the relative positions of capacitor  304  and resistor  305  is swapped, then the coupled RF energy at capacitive element  304  will be proportional to an incident wave propagating from right-to-left (Reflected or reverse power detection). For both forward and reverse power detection applications, the basic geometric constraint for the helix are the same. 
     As illustrated in  FIG. 3 , a series of vias  303  are employed to interconnect the upper and lower plates  300  and  302  to provide electrical connectivity to the entire helix coupling structure. The distance from the vias to the edge of the main transmission line is determined by the plane of intersection between the H-field and the helix structure. Although this relationship will be explored in further detail herein, it should be noted that in some alternative embodiment geometries, the via connection  303  can be considered perpendicular “flanges” in the Z-plane to the H-field vectors in the XY-plane (vias  303  perpendicular to H-field of transmission line  301 ). This will be explored further in subsequent descriptions. 
     In general, all multi-layered helix structures must be interconnected to form the proper geometry, which in turn must be attached to a terminating impedance and to a series capacitive element (lump or distributed) as shown in  FIG. 3 . The frequency response of the coupler is determined by the intrinsic inductance of the helix, and the series lump element capacitor. Directivity is “peak” in the frequency band of operation, providing optimum performance. The series capacitor also may serve as a dc block capacitor for external transmitter circuits that require a particular bias. 
     Referring to  FIG. 4 , the interrelationship between the helix structure, secondary flanges, and main transmission line  401  can be clearly seen. This top view of  FIG. 3  shows that the horizontal separation  408  between the flange edge and the transmission line  401  is minimal without overlapping each other. The distance  404  from the transmission line edge to vias  403  is significantly larger than flange separation  408 . And since near field coupling efficiency is inversely proportional to the distance between the source (transmission line  401 ) and receptor, the flanges embedded into the upper and lower plates  400  and  402  are the primary coupling structure over the vias. The flange segment lengths  405 ,  406 , and  407  are determined by geometric spacing of the interconnecting vias  409  and the frequency of operation. 
     It should be noted that in this embodiment, the flanges are variable length to maximize coupling bandwidth and that the upper and lower plate flanges do not overlap each other. It will be evident to those skilled in the art, that variable number of flanges (greater than one) can be used and the flange length  405 ,  406 , and  407  can be varied from this embodiment and still functionally maintain the integrity of the helix structure. The dielectric thickness between the planes containing lower plates  402 , transmission line  401  and upper plates  400  are determined by standard PCB manufacturing process and desired coupling efficiency, understanding that thinner dielectric thickness (closer proximity between transmission line and coupling plates) increase coupling efficiency for reasons previously discussed. The illustration of the multi-layered structure is shown in  FIG. 7 . This coupling structure provides H-field coupling in the XY plane primarily through the flange extensions. As previously discussed, capacitive and resistive terminations may be positioned at terminals  3  and  4  to functionally differentiate the direction (incident or reflected) of the RF energy being conducted on transmission line  401 . 
     Referring to  FIG. 5 , a second or alternative embodiment of the invention is disclosed. Here, no parallel structures are evident in the XY plane between transmission line  501  and upper and lower plate structures  500  and  502 . However, the transmission line  501  is offset to one side to provide close proximity to the family of vias  504 . Distance  505  is significantly smaller than distance  504  (proximity to family of vias  506 ). Thus the H-field radiation in the XY plane is perpendicularly bisected by vias  503  in the Z-plane. Distance  505  is set by PCB manufacturing tolerances and the desired level of coupling efficiency required for the given application. The geometry in  FIG. 5  has a further advantage of minimizing capacitive coupling by minimizing the overlap surface area between coupling plates  500  and  502  and transmission line  501 . The distance  509  between each via is determined by the desired arch-length of the interconnecting coupling plates  500  and  502 , and the desired coupling coefficient. The greater the number of vias per unit length of transmission line  501 , the higher the coupling factor. 
     It should be further noted that distance  504  should be large compared to distance  505 , to avoid degradation of coupling efficiency and directivity. This degradation results from opposing H-Field induce in the upper and lower plate  500  and  502  that meet at via connection  506 . As distance  504  is reduced, the upper and lower plate region at vias  506  tend to become parallel to transmission line  501 , increasing the current density J induced on each plate by the H-Field. However, the H-field vector at the bottom of upper plate  500  opposes the H-field vector at the top lower plate  502 . These two opposing fields are electrically connected together at via  506 , thereby reducing the coupling efficiency. This is clearly illustrated in  FIG. 2  while visualizing the upper and lower plates  200  and  202  connected end-to-end through a series of vias. This condition does not exist at via connections  503 , as the upper and lower plate H-fields are additive, and are also located at the point of highest current density (closest proximity to transmission line  501 ). 
     In yet another embodiment shown in  FIG. 6 , a coupling structure contains flange structures “embedded” into the upper and lower plates  600  and  602 .  FIG. 6  illustrates a helix coupler that bisects the H-field in the XY plane utilizing unique plate geometries to provide maximal surface area parallel to the transmission line  601  (orthogonal to the H-field). As in the previous embodiments, the upper and lower plates are interconnected through vias  603  so as to create a overall helix structure around the main transmission line  601 . This coupler design is a variation of the previously discussed “yag coupler” shown in  FIG. 4 . The modification in this embodiment eliminates the “open stub” effect in the Yag-coupler, which creates high voltage densities at the open end of the flange with no connectivity to provide for current flow. By minimizing the open stub in the coupler structure, the overall coupler resistivity as seen by the induced current is reduced. This improves coupling efficiency and directivity of the coupler structure. 
     As in the previous embodiments, the distance  609  between vias is determined by manufacturing tolerances, desired coupling efficiency, and frequency of operation (which sets the number of turns in the helix). The width of transmission line  601  is determined using classical stripline or transmission line calculations dependent on substrate dielectric constant, desired characteristic impedance, and distance to nearest ground planes. The substrate geometry is equivalent to the previously discussed embodiments and is illustrated in  FIG. 7 . The length  604 ,  605  and  606  of each parallel segment embedded within the upper and lower plates  600  and  602  is set by via separation  609  and angle of intersection  607 . It should be apparent to those skilled in the art that variations of the plate geometry can be achieved that changing the relative angle of  607  so that the lengths  604 ,  605  and  606  are varied, while still maintaining the integrity of the proposed embodiment. 
     Another possible embodiment of this invention is applicable to non-planar structures. Referring to  FIG. 8 , a transmission line or wire  801  is suspended in a dielectric such as air. A semi-rigid, continuous helix structure  800  is wrapped around transmission line  801  to support secondary coupling element  802 . These coupling elements are parallel to the transmission line and may be of varying lengths  804 , varying quantity, or separation  803 , depending on the application, frequency of operation, or desired performance. The coupling elements  802  are connected together through helix structure  800 , which is electrically conducting. Termination elements such as capacitor  806  and resistor  805  are positioned at the terminals of the helix to provide directional differentiation of the RF energy traveling along transmission line  801 . This is similar to the techniques described in the previous embodiments. 
     While the preferred embodiments of the invention have been illustrated and described, it will be clear that the invention is not so limited. Numerous modifications incorporating combinations of the disclosed embodiments, changes, or variations, substitutions and equivalents of other geometries will occur to those skilled in the art without departing from the spirit and scope of the present invention as defined by the appended claims.