Abstract:
A phase comparator includes a first signal input terminal supplied with a reference signal, a second signal input terminal supplied with an input signal to be compared with the reference signal, a first gate circuit having a pair of input terminals and an output terminal, one of the pair of input terminals of which is connected to the first signal input terminal, a second gate circuit having a pair of input terminals and an output terminal, one of the pair of input terminals of which is connected to the second signal input terminal, a first bi-stable circuit having set, reset and output terminals, the set terminal of which is connected to the output terminal of the first gate circuit, a second bi-stable circuit having set, reset and output terminals, the set terminal of which is connected to the output terminal of the second gate circuit, and first and second gate control circuits connected between the other input terminals of the first and second gate circuits and the output terminals of the first and second bi-stable circuits operative to open one of the first and second gate circuit alternately. The output signal proportional to the phase difference between the input signal and the reference signal is produced at one of the output terminals of the first and second bi-stable circuit in accordance with whether the input signal phase-advances more than the reference signal or not.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates generally to a phase comparator and is directed more particularly to a phase comparator for use in a phase locked loop circuit. 
     2. Description of the Prior Art 
     In general, a phase locked loop circuit (hereinafter, simply referred to as a PLL circuit) includes a voltage-controlled oscillator, a phase comparator and so on. The phase comparator is supplied with, for example, a reference signal and an oscillating signal from a voltage-controlled oscillator, in which they are compared for phase and then this phase comparator produces an error signal in accordance with a phase difference between these two signals. Then, this error signal is supplied through, for example, a low pass filter to the voltage-controlled oscillator as a control signal, so that the output signal from this voltage-controlled oscillator is synchronized with the reference signal. 
     In the past, as the phase comparator utilized in such PLL circuit, there has been proposed such one as illustrated in FIG. 1. 
     In FIG. 1, reference numeral 1a generally denotes an input terminal supplied with, for example, a reference signal f 1  and 1b denotes an input terminal to which an oscillating signal f 2  is supplied from, for example, a voltage-controlled oscillator. Numerals 2 to 10 respectively designate NAND circuits in which each pair of the NAND circuits 2 and 9, 3 and 4, 5 and 6 and 7 and 10 forms a flip-flop circuit. In addition, numerals 11a and 11b represent one and the other output terminals at which signals corresponding to the phase difference between the aforesaid reference signal f 1  and oscillating signal f 2  are produced. 
     In the phase comparator thus composed, as shown in FIGS. 2A and 2B, when the frequency of the oscillating signal f 2  which will be supplied to the input terminal 1b is lower than that of the reference signal f 1  which will be supplied to the input terminal 1a, at the one and other output terminals 11a and 11b are produced such signals S 01  and S 02  as shown in FIGS. 2C and 2D. That is, at the output terminal 11a is achieved the signal S 01  which has a longer period of low level &#34;0&#34; if the frequency of the oscillating signal f 2  is lower and lower as compared with the frequency of the reference signal f 1 , while at the output terminal 11b is produced the signal S 02  which is always at high level &#34;1&#34;. Accordingly, by supplying these output signals S 01  and S 02  to, for example, the voltage-controlled oscillator as the control signal, it is possible to control the voltage-controlled oscillator such that the frequency and the phase of the oscillating signal f 2  may take the same direction as those of the reference signal f 1 . 
     On the other hand, as shown, for example, in FIGS. 3A and 3B, when the frequency of the oscillating signal f 2  which will be supplied to the input terminal 1b is same as that of the reference signal f 1  which will be supplied to the input terminal 1a but the phase thereof is delayed from the reference signal f 1  by φ, at the one and other output terminals 11a and 11b are derived such signals S 01  and S 02  as shown in FIGS. 3C and 3D. That is, at the one output terminal 11a is achieved the signal S 01  which becomes the low level &#34;0&#34; periodically only in the period responsive to the phase difference φ, whereas at the other output terminal 11b is achieved the signal S 02  which always becomes the high level &#34;1&#34;. Therefore, by supplying these output signals S 01  and S 02  to, for example, the voltage-controlled oscillator as the control signal, it is possible to control the voltage-controlled oscillator such that the phase of the oscillating signal f 2  may take the same direction as that of the reference signal f 1 . 
     As described above, according to the phase comparator as illustrated in FIG. 1, in accordance with the phase difference of the oscillating signal f 2  relative to the reference signal f 1 , at the one and other output terminals 11a and 11b can be derived the signals S 01  and S 02  each corresponding thereto. Thus, by controlling, for example, the voltage-controlled oscillator in association with these output signals S 01  and S 02 , it is possible to coincide the frequency and the phase of the oscillating or comparing signal f 2  with those of the reference signal f 1 . 
     Nevertheless, such phase comparator as seen in FIG. 1 utilizes four flip-flop circuits, each comprised of the NAND circuits (pairs of the NAND circuits 2 and 9, 3 and 4, 5 and 6, 7 and 10 compose the flip-flop circuits, respectively), so that the wirings thereamong are very complicated, and in relation to the number of circuit elements or logics used therein, the conventional phase comparator is relatively expensive. 
     OBJECTS AND SUMMARY OF THE INVENTION 
     Therefore, it is an object of the present invention to provide a phase comparator which can obviate the aforesaid defects inherent to the prior art. 
     It is another object of the present invention to provide a phase comparator particularly suitable for use in a phase locked loop circuit. 
     In accordance with the aspects of the invention, a phase comparator is disclosed, which includes a first signal input terminal supplied with a reference signal, a second signal input terminal supplied with an input signal to be compared with the reference signal, a first gate circuit having a pair of input terminals and an output terminal, one of the pair of the input terminals of which is connected to the first signal input terminal, a second gate circuit having a pair of input terminals and an output terminal, one of the pair of input terminals of which is connected to the second signal input terminal, a first bi-stable circuit having set, reset and output terminals, the set terminal of which is connected to the output terminal of the first gate circuit, a second bi-stable circuit having set, reset and output terminals, the set terminal of which is connected to the output terminal of the second gate circuit, a first gate control device connected between the output terminal of the second bi-stable circuit and the other of the pair of the input terminals of the first gate circuit, a second gate control device connected between the output terminal of the first bi-stable circuit and the other of the pair of the input terminals of the second gate circuit, a first resetting device connected between the first signal input terminal and the reset terminal of the second bi-stable circuit, and a second resetting device connected between the second signal input terminal and the reset terminal of the first bi-stable circuit. 
     The other objects, features and advantages of the present invention will become apparent from the following description taken in conjunction with the attached drawings. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a circuit block diagram schematically showing one example of a conventional phase comparator; 
     FIGS. 2A to 2D and FIGS. 3A to 3D are respectively waveform diagrams used to explain the operation of the prior art phase comparator seen in FIG. 1; 
     FIGS. 4 and 5 are state transition diagrams each used to explain the stable point of the conventional phase comparator as described above in FIG. 1; 
     FIG. 6 is a fundamental state transition diagram showing a stable point of a phase comparator which is utilized in the present invention; 
     FIG. 7 is a circuit block diagram showing one embodiment of the phase comparator according to the present invention; 
     FIGS. 8A to 8F are waveform diagrams used to explain the operation of a trigger pulse oscillator used in the present invention; 
     FIG. 9 is a state transition diagram showing a practical operation of the phase comparator according to the present invention; 
     FIGS. 10A to 10D and FIGS. 11A to 11J are waveform diagrams used to explain the operation of the phase comparator according to the present invention; and 
     FIGS. 12 to 18 are circuit block diagrams each showing other embodiments of the phase comparator according to the present invention. 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     The embodiments of the phase comparator according to the present invention will hereinafter be described with reference to the drawings. 
     By the way, FIG. 4 is a state transition diagram showing the state transition of the prior art phase comparator previously discussed with reference to FIG. 1. As will be apparent from FIG. 4, through the experiments, it was ascertained that the conventional phase comparator was transited with eight stable states of U 1 , U 2 , O 1 , O 2 , O 3 , O 4 , D 1  and D 2 . 
     In this case, the signals S 01  and S 02  produced at the output terminals 11a and 11b (FIG. 1) become &#34;0&#34; and &#34;1&#34; when the prior art phase comparator is in the states of U 1  and U 2 , &#34;1&#34; and &#34;1&#34; when O 1 , O 2 , O 3  and O 4 , and &#34;1&#34; and &#34;0&#34; when D 1  and D 2 , respectively. Accordingly, it may be considered that these eight stable states of U 1 , U 2 , O 1 , O 2 , O 3 , O 4 , D 1  and D 2  will be collected into, for example, three sets as shown in FIG. 5. 
     To develop the above consideration further, as shown by the state transition diagram in FIG. 6, if three stable states U, O and D (in the state U, S 01  =&#34;0&#34; and S 02  =&#34;1&#34;, in the state O, S 01  =&#34;1&#34; and S 02  =&#34;1&#34; and in the state D, S 01  =&#34;1&#34; and S 02  =&#34;0&#34;) are taken for a phase comparator and if the phase comparator is changed in state along the rule such as shown in the figure, it was ascertained that output signals same as those of the prior art phase comparator described above could be obtained. 
     Therefore, one embodiment of a phase comparator according to the present invention for achieving the aforesaid transition characteristic in FIG. 6 will hereinafter be described with reference to FIG. 7. In FIG. 7, the like references corresponding to those of FIG. 1 designate the same elements and parts and hence they will not be described in detail for simplicity. 
     As illustrated in FIG. 7, the input terminal 1 a  which will be supplied with, for example, the reference signal f 1  is connected to one input terminal or side of a NOR circuit 13 a1  comprising a trigger pulse generator or oscillator 13 a , and also connected through an inverter 13 a2  to the other input side of the NOR circuit 13 a1 . In this case, a delay time of the inverter 13 a2  is selected as 2τ. By way of example, if an input signal f 1  such as, shown in FIG. 8A is supplied to the input terminal 1 a , the inverter 13 a2  produces at its output side a signal such as, shown in FIG. 8B. Accordingly, the NOR circuit 13 a1  produces at its output terminal a trigger pulse having a pulse width 2τ as shown in FIG. 8C when the signal which is supplied to the input terminal 1 a  falls down. 
     The output side of this NOR circuit 13 a1  is connected to one input side of an AND circuit 14 a  forming a gate circuit and the output side of this AND circuit 14 a  is connected to a set signal input terminal S a  of a flip-flop circuit 15 a . 
     The input terminal 1 b  to which a signal such as, the oscillating signal f 2  is supplied from the voltage-controlled oscillator, is connected to one input side of a NOR circuit 13 b1  forming a trigger pulse generator or oscillator 13 b  and also connected through an inverter 13 b2  to the other input side of the NOR circuit 13 b1 . This trigger pulse oscillator 13 b  is constructed same as the trigger pulse oscillator 13 a  as described above, in which as the output of the NOR circuit 13 b1  is produced a trigger pulse having a pulse width 2τ at the falling-down edge of the signal which is supplied to the input terminal 1 b . 
     The output side of this NOR circuit 13 b1  is connected to one input side of an AND circuit 14 b  comprising a gate circuit and the output side of this AND circuit 14 b  is connected to a set signal input terminal S b  of a flip-flop circuit 15 b . 
     Also, the output side of the NOR circuit 13 b1  is connected to a reset signal input terminal R a  of the flip-flop circuit 15 a  and an inverted output terminal Qa of this flip-flop circuit 15 a  is connected to the other input side of the AND circuit 14 b . 
     The output side of the NOR circuit 13 a1  is also connected to a reset signal input terminal R b  of the flip-flop circuit 15 b  and an inverted output terminal Qb of this flip-flop circuit 15 b  is connected to the other input side of the AND circuit 14 a . 
     Then, from the output terminals Qa and Qb of the flip-flop circuits 15 a  and 15 b  are led out the one and other output terminals 11 a  and 11 b . 
     In this case, the trigger pulse generated from the trigger pulse oscillator 13 a  is supplied through the AND circuit 14 a  to the set signal input terminal Sa of the flip-flop circuit 15 a , whereby this flip-flop circuit 15 a  is set. That is, the flip-flop circuit 15 a  is set in such a state that its output terminal Qa will produce the high level signal &#34;1&#34; and its inverted output terminal Qa will produce the low level signal &#34;0&#34;. In this case, however, when the flip-flop circuit 15 b  is in the set state, that is, when the flip-flop circuit 15 b  is in the state such that the low level signal &#34;0&#34; is produced at the inverted output terminal Qb, the gate circuit formed of the AND circuit 14a is in the open state. Thus the trigger pulse from the trigger pulse oscillator 13 a  is not supplied to the set signal input terminal Sa of the flip-flop circuit 15 a , so that this flip-flop circuit 15 a  is not set. In other words, the set of the flip-flop circuit 15 a  is inhibited. 
     The trigger pulse derived from the trigger pulse oscillator 13 a  is also supplied to the reset signal input terminal R b  of the flip-flop circuit 15 b , by which the flip-flop circuit 15 b  is reset. That is, the flip-flop circuit 15 b  is made in the state such that the low level signal &#34;0&#34; is produced at its output terminal Qb and the high level signal &#34;1&#34; is produced at the inverted output terminal Qb thereof. 
     Similarly, the trigger pulse produced from the trigger pulse oscillator 13 b  is delivered through the AND circuit 14 b  to the set signal input terminal Sb of the flip-flop circuit 15 b , by which this flip-flop circuit 15 b  is set. That is, the flip-flop circuit 15 b  is made in the state such that the high level signal &#34;1&#34; is produced at the output terminal Qb and the low level signal &#34;0&#34; is produced at the inverted output terminal Qb. However, also in this case, when the flip-flop circuit 15 a  is in the set state, that is, when the low level signal &#34;0&#34; is produced at the inverted output terminal Qa thereof, the gate circuit comprised of the AND circuit 14 b  is in the open state. Thus the trigger pulse from the trigger pulse oscillator 13 b  is not supplied to the set signal input terminal Sb of the flip-flop circuit 15 b , so that this flip-flop circuit 15 b  is not set. In other words, the set of the flip-flop circuit 15 b  is inhibited. 
     Further, the trigger pulse generated from the trigger pulse oscillator 13 b  is delivered to the reset signal input terminal Ra of the flip-flop circuit 15 a  to thereby permit the flip-flop circuit 15 a  to be reset. That is, the flip-flop circuit 15 a  is made in the state that the low level signal &#34;0&#34; is produced at the output terminal Qa and the high level signal &#34;1&#34; is produced at the inverted output terminal Qb. 
     In addition, in this case, when the trigger pulses to set and reset these flip-flop circuits 15 a  and 15 b  are supplied together thereto at the same time, because of the delay by the AND circuits 14 a  and 14 b , the reset is dominant to the set so that these flip-flop circuits 15 a  and 15 b  are made in the reset states. 
     In this way, according to the phase comparator of this embodiment shown in FIG. 7, when the signal f 1  which is supplied to the input terminal 1a falls down (that is, it is changed from the high level &#34;1&#34; to the low level &#34;0&#34;), the flip-flop circuits 15 a  and 15 b  are set and reset, respectively. Whereas, when the signal f 2  which is supplied to the input terminal 1b falls down, the flip-flop circuits 15 a  and 15 b  are reset and set, respectively. However, in this case, if the one of the flip-flop circuits 15 a  and 15 b  is already in the set state, the other is inhibited from being set. 
     FIG. 9 shows a state transition of the phase comparator according to this embodiment which has three stable states U&#39;, O&#39; and D&#39; (in the state of U&#39;, S 01  =&#34;1&#34; and S 02  =&#34;0&#34;, in the state of O&#39;, S 01  =&#34;0&#34; and S 02  =&#34;0&#34; and in the state of D&#39;, S 01  =&#34;0&#34; and S 02  =&#34;1&#34;) and which is transited in state along the rule as shown in the figure. This state transition diagram of FIG. 9 is equivalent to the simplified state transition diagram of the phase comparator as shown in FIG. 6. 
     Therefore, the phase comparator of this embodiment presents the state transition nearly same as that of the phase comparator with the state transition shown in FIG. 6. 
     When the phase comparator of this embodiment is supplied at its input terminals 1 a  and 1 b  with the signals f 1  and f 2  (f 2  &gt;f 1  in frequency) such as shown in FIGS. 10A and 10B, the signals S 01  and S 02  derived at the one and other output terminals 11 a  and 11 b  thereof become those shown in FIGS. 10C and 10D, respectively. That is, at the output terminal 11 b  is produced the signal S 02  whose period of the low level &#34;0&#34;  changes in accordance with the difference between the frequency of the oscillating signal f 2  and that of the reference signal f 1 , and at the output terminal 11 a  is produced the signal S 01  which is always the low level &#34;0&#34;. Accordingly, if these output signals S 01  and S 02  are supplied to, for example, the voltage-controlled oscillator as the control signal, it is possible to control the voltage-controlled oscillator such that the frequency and the phase of the oscillating signal f 2  may become the same as those of the reference signal f 1 . Although in this example, the initial values of the output signals S 01  and S 02  are respectively &#34;0&#34; and &#34;0&#34;, in other cases, that is, even when the initial values of the signals S 01  and S 02  are &#34;0&#34; and &#34;1&#34;, &#34;1&#34; and &#34;0&#34;, and &#34;1&#34; and &#34;1&#34;, respectively, since the initial values of the output signals S 01  and S 02  are reset at a time point t 1 , thereafter they become same as one other. 
     Also, when the phase comparator according to this embodiment of FIG. 7 is supplied at its input terminals 1 a  and 1 b  with the signals f 1  and f 2  such as shown in FIGS. 11A and 11B (signals f 1  and f 2  are same one another in frequency but have a phase difference φ&#39; therebetween), the signals S 01  and S 02  produced at the one and other output terminals 11 a  and 11 b  become respectively those shown in FIGS. 11C and 11D. That is, at the one output terminal 11 a  is produced the signal S 01  which periodically becomes high level &#34;1&#34; during only the period corresponding to the phase difference φ&#39; and at the other output terminal 11 b  is provided the signal S 02  which always becomes the low level &#34;0&#34;. Consequently, if these output signals S 01  and S 02  are supplied to, for example, the voltage-controlled oscillator as the control signal, it is possible to control the voltage-controlled oscillator such that the phase of the oscillating signal f 2  will become the same as that of the reference signal f 1 . While FIG. 11C and 11D are the cases where the initial values of these output signals S 01  and S 02  are &#34;0&#34; and &#34;0&#34;, when the initial values of the output signals S 01  and S 02  are &#34;0&#34; and &#34;1&#34;, these output signals S 01  and S 02  become those shown in FIGS. 11E and 11F, when &#34;1&#34; and &#34;0&#34;, they become those shown in FIGS. 11G and 11H and when &#34;1&#34; and &#34;1&#34;, they become those shown in FIGS. 11I and 11J, respectively. 
     As described above, the phase comparator according to the present invention has by no means an inferior function as compared with the conventional phase comparator. In addition, since it is formed of two flip-flop circuits 15 a  and 15 b , the circuit arrangement thereof is very simple and hence it becomes less expensive as compared with the conventional phase comparator. 
     Next, with reference to FIGS. 12 through FIGS. 18, other embodiments of the phase comparator according to the present invention will be described. In FIGS. 12 to 18, the like parts corresponding to those of FIG. 7 are marked with the same references and their detailed explanation will be omitted. 
     In the phase comparator shown in FIG. 12, the output side of the inverter 13 a2  is connected to the reset signal input terminal Rb of the flip-flop circuit 15 b  and the output side of the inverter 13 b2  is connected to the reset signal input terminal Ra of the flip-flop circuit 15 a . The other circuit construction is substantially arranged same as that of the embodiment in FIG. 7. 
     The phase comparator shown in FIG. 12 has the similar operation and effect to those of the embodiment of FIG. 7 and in addition, the flip-flop circuits 15 a  and 15 b  are reset by the signal as shown in FIG. 8B, so that for example, when the signals f 1  and f 2  each supplied to the input terminals 1 a  and 1 b  fall down at the same time, both of the flip-flop circuits 15 a  and 15 b  can be reset surely. 
     Further, in the phase comparator as shown in FIG. 13, there are further provided trigger pulse generators 16 a  and 16 b  and the flip-flop circuits 15 a  and 15 b  are respectively reset by the trigger pulses produced from these trigger pulse oscillators 16 a  and 16 b . In this case, a delay time which will be decided by inverters 16 a2  and 16 b2  comprising the respective trigger pulse oscillators 16 a  and 16 b  is selected as 3τ. If the input terminals 1 a  and 1 b  are supplied with the signal as shown in FIG. 8A, the inverters 16 a2  and 16 b2  produce at their output sides the signal such as shown FIG. 8D, so that NOR circuits 16 a1  and 16 b1  produce at their output sides trigger pulses each having a pulse width 3τ at the falling-down of the signal supplied to the input terminals 1 a  and 1 b  as shown in FIG. 8E. Then, these trigger pulses are supplied to the reset signal input terminals R.sub. b and R a  of the flip-flop circuits 15 b  and 15 a . The other elements and parts are constructed same as those of the embodiment in FIG. 7. 
     Next, in the phase comparator as illustrated in FIG. 14, as the NOR circuits 13 a1 , and 13 b1  forming the trigger pulse oscillators 13 a  and 13 b , are utilized three-input NOR circuits, by which the gate circuits for use in inhibiting the flip-flop circuits 15 a  and 15 b  from being set are constructed. 
     In the embodiments of the phase comparators shown in FIGS. 13 and 14, it is possible to achieve the same operation and effect as those of the phase comparator as seen in FIG. 12. 
     Next, FIG. 15 shows a further example of the phase comparator according to the invention in which a four-input NOR circuit 17 constructs the trigger pulse generator and the gate circuits for use in inhibiting the flip-flop circuits 15 a  and 15 b  from being set. 
     Further, in the phase comparator of the invention shown in FIG. 16, there are utilized three-input AND circuits 13 a3  and 13 b3  which provide the trigger pulse oscillators 13 a  and 13 b  and the gate circuits for use in inhibiting the flip-flop circuits 15 a  and 15 b  from being set. In this case, when a delay time by the inverters 13 a2  and 13 b2  is specified as 2τ, these AND circuits 13 a3  and 13 b3  produce at their output sides the trigger pulses each having the pulse width 2τ at the rising-up of the signals supplied to the input terminals 1 a  and 1 b  as shown in FIG. 8F, which are respectively supplied to the set signal input terminals Sa and Sb of the flip-flop circuits 15 a  and 15 b . 
     Also in the embodiments of the phase comparators shown in FIGS. 15 and 16, it is possible to achieve the same operation and effect as those of the embodiment in FIG. 7. 
     Further, FIG. 17 shows another embodiment of the phase comparator according to the invention in which trigger pulse generators 18 a  and 18 b  are further provided for the phase comparator seen in FIG. 16 and the trigger pulses formed from these trigger pulse oscillators 18 a  and 18 b  are supplied to the reset signal input terminals R b  and R a  of the flip-flop circuits 15 b  and 15 a . 
     In addition, FIG. 18 shows a yet further embodiment of the phase comparator of the invention in which the trigger pulse oscillators 18 a  and 18 b  are also provided for the phase comparator seen in FIG. 15 and the trigger pulses formed from these trigger pulse oscillators 18 a  and 18 b  are supplied to the reset signal input terminals R b  and R a  of the flip-flop circuits 15 a  and 15 b . With such embodiments as seen in FIGS. 17 and 18, it is also possible to achieve the same operation and effects as those of the embodiments shown in FIGS. 16 and 15. In addition, the phase comparators shown in FIGS. 17 and 18 can operate satisfactorily irrespective of the waveforms of the input signals f 1  and f 2  supplied to the input terminals 1 a  and 1 b . By the way, in the case of the embodiments seen in FIGS. 15 and 16, the waveforms of the input signals f 1  and f 2  have to be less than 50% in duty ratio. 
     In the above embodiments of the invention, while the flip-flop circuits 15 a  and 15 b  and so on are constructed by using the NOR circuits, instead of these NOR circuits, the NAND circuits may be utilized to construct them. If so, only the phase of the output signals S 01  and S 02  are inverted, which causes no inconvenience or disadvantage at all. 
     The above description is given on the preferred embodiments of the invention, but it will be apparent that many modifications and variations could be effected by one skilled in the art without departing from the spirits or scope of the novel concepts of the invention, so that the scope of the invention should be determined by the appended claims only.