Abstract:
Path delays experienced by a received signal are estimated for use in, for example, a RAKE receiver. This involves hypothesizing a plurality of path delays. For each of a plurality of measurement time slots and for each of the hypothesized path delays, a measurement based on the received signal is made along with a determination of whether a fade occurred. For each of the plurality of measurement time slots and for each of the hypothesized path delays, the measurement is combined with a corresponding one of a plurality of cumulative metrics only if it was determined that no fade occurred. Then, for each of the plurality of hypothesized path delays, the corresponding one of the plurality of cumulative metrics is used to determine whether the hypothesized path delay corresponds to a real path delay.

Description:
BACKGROUND  
         [0001]    The present invention relates to communications systems, more particularly to RAKE receivers, and even more particularly to techniques for estimating path delays associated with a communications channel.  
           [0002]    A cellular communication system is one in which a large geographical area is divided up into so-called cells. Each cell is served by a corresponding base station, which uses radiocommunication techniques to link mobile units located within the cell to a land-based part of the cellular communication system. The land-based part of the cellular communication system is capable not only of linking communications between mobile units located in the same or different cells, but also of connecting a mobile user to other communications networks, such as a Public Switched Telephone Network (PSTN) and/or a computer-oriented data network. In this way, a mobile user may be capable of establishing a call to (or receiving a call from) a non-mobile telephone (a so-called “Plain Old Telephone”, or “POT”) or to processing equipment connected to a computer network. When a user of a mobile unit moves from one cell to another, responsibility for maintaining any ongoing call that he may be participating in shifts from the original cell to the new “target” cell, in an operation referred to as “handoff” or “handover”.  
           [0003]    The cellular radiocommunication industry has made phenomenal strides in commercial operations in the United States as well as the rest of the world. Growth in major metropolitan areas has far exceeded expectations and is rapidly outstripping system capacity. If this trend continues, the effects of this industry&#39;s growth will soon reach even the smallest markets. Innovative solutions are required to meet these increasing capacity needs as well as to maintain high quality service and avoid rising prices.  
           [0004]    Throughout the world, one important step in the advancement of radio communication systems is the change from analog to digital transmission. Equally significant is the choice of an effective digital transmission scheme for implementing next generation technology. Furthermore, it is widely believed that the first generation of Personal Communication Networks (PCNs), employing low cost, pocket-sized, cordless telephones that can be carried comfortably and used to make or receive calls in the home, office, street, car, and the like, will be provided by, for example, cellular carriers using the next generation digital cellular system infrastructure. An important feature desired in these new systems is increased traffic capacity, and efficient use of this capacity.  
           [0005]    Currently, channel access is very often achieved using Frequency Division Multiple Access (FDMA) and Time Division Multiple Access (TDMA) methods. In FDMA, a communication channel is a single radio frequency band into which a signal&#39;s transmission power is concentrated. Signals that can interfere with a communication channel include those transmitted on adjacent channels (adjacent channel interference) and those transmitted on the same channel in other cells (co-channel interference). Interference with adjacent channels is limited by the use of band pass filters which only pass signal energy within the specified frequency band. Co-channel interference is reduced to tolerable levels by restricting channel reuse such that a minimum separation distance is required to exist between cells in which the same frequency channel is used. Thus, with each channel being assigned a different frequency, system capacity is limited by the available frequencies as well as by limitations imposed by channel reuse.  
           [0006]    In TDMA systems, a channel consists of, for example, a time slot in a periodic train of time intervals over the same frequency. Each period of time slots is called a frame. A given signal&#39;s energy is confined to one of these time slots on a given frequency. Adjacent channel interference is limited by the use of a time gate or other synchronization element that only passes signal energy received at the proper time. Thus, with each channel being assigned a different time slot, system capacity is limited by the number of available time slots per frequency, as well as by limitations imposed by channel reuse as described above with respect to FDMA, which determines the number of frequencies that are available for use in any given cell.  
           [0007]    With FDMA and TDMA systems (as well as hybrid FDMA/TDMA systems), one goal of system designers is to ensure that two potentially interfering signals do not occupy the same frequency at the same time. In contrast, Code Division Multiple Access (CDMA) is a channel access technique that allows signals to overlap in both time and frequency. CDMA is a type of spread spectrum communication technique, which has been around since the days of World War II. Early applications were predominantly military oriented. However, today there has been an increasing interest in using spread spectrum systems in commercial applications because spread spectrum communications provide robustness against interference, which allows for multiple signals to occupy the same bandwidth at the same time. Examples of such commercial applications include digital cellular radio, land mobile radio, and indoor and outdoor personal communication networks.  
           [0008]    In a CDMA system, each signal is transmitted using any of a number of spread spectrum techniques. In some variations of CDMA, the informational data stream to be transmitted is impressed upon a much higher rate data stream known as a signature sequence. Typically, the signature sequence data are binary, thereby providing a bit stream. One way to generate this signature sequence is with a pseudo-noise (PN) process that appears random, but can be replicated by an authorized receiver. The informational data stream and the high bit rate signature sequence stream are combined by multiplying the two bit streams together, assuming the binary values of the two bit streams are represented by +1 or −1. This combination of the higher bit rate signal with the lower bit rate data stream is called spreading the informational data stream signal. Each informational data stream or channel is allocated a unique signature sequence.  
           [0009]    A plurality of spread information signals modulate a radio frequency carrier, for example by Binary Phase Shift Keying (BPSK), and are jointly received as a composite signal at the receiver. Each of the spread signals overlaps all of the other spread signals, as well as noise-related signals, in both frequency and time. If the receiver is authorized, then the composite signal is correlated with one of the unique signature sequences, and the corresponding information signal can be isolated and despread. If quadrature phase shift keying (QPSK) modulation is used, then the signature sequence may consist of complex numbers (having real and imaginary parts), where the real and imaginary parts are used to modulate respective ones of two carriers at the same frequency, but ninety degrees out of phase with respect to one another.  
           [0010]    Traditionally, for example in the case of BPSK modulation, a signature sequence is used to represent one bit of information. Receiving the transmitted sequence or its complement indicates whether the information bit is a +1 or −1, sometimes denoted “0” or “1”. The signature sequence usually comprises N bits, and each bit of the signature sequence is called a “chip”. The entire N-chip sequence, or its complement, is referred to as a transmitted symbol. The conventional receiver, such as a RAKE receiver, correlates the received signal with the complex conjugate of the known signature sequence to produce a correlation value. Only the real part of the correlation value is computed. When a large positive correlation results, a “0”is detected; when a large negative correlation results, a “1” is detected.  
           [0011]    RAKE receivers are well-known. A conventional baseband processor  100  in a typical coherent RAKE receiver is illustrated in FIG. 1. The baseband signal is provided to a bank of correlators  101 , which correlate different delays of the received signal to the despreading code, producing despread values. The delays are provided by channel delay estimator  103 , which uses known methods to estimate the delays, such as finding delays which give large despread values. The despread values corresponding to different delays are combined in combiner  105  using a weighted sum. The weights are the conjugates of channel coefficient estimates provided by channel coefficient estimator  107 . For example, correlations to a pilot signal can be used to obtain channel coefficients.  
           [0012]    Consider a simple example in which the received chip-spaced baseband samples during one symbol period is represented by r(k). These samples are modeled as  
             r ( k )= bc   0   s ( k )+ bc   1   s ( k− 1)+ w ( k )  (1)  
           [0013]    where b is the symbol sent, c 0  and c 1  are the channel coefficients, the delays are 0 and 1 chip period, s(k) is the chip sequence used to spread the symbol, and w(k) are impairment (noise plus interference) samples.  
           [0014]    The bank of correlators  101  produces two despread values, denoted x 0  and x 1 , corresponding to the two rays. These can be expressed as  
               x   0     =       1   L            ∑     k   =   0       L   -   1                           s   *          (   k   )            r        (   k   )                     (   2   )                 x   1     =       1   L            ∑     k   =   0       L   -   1                           s   *          (   k   )            r        (     k   +   1     )                     (   3   )                               
 
           [0015]    where the superscript “*” denotes complex conjugation and L is the despreading factor. Division by L is shown for illustrative purposes, while in practice it is well known how to extend results to the case in which the division is omitted.  
           [0016]    The combiner  105  combines the two despread values using estimates of the channel coefficients denoted ĉ 0  and ĉ 1 , to produce a detection statistic that corresponds to an information symbol. This can be expressed as  
             z×ĉ   0   *x   0   +ĉ   1   *x   1   (4)  
           [0017]    The symbol value that is closest to z gives the detected value {circumflex over (b)}. For BPSK modulation, b is either +1 or −1, so that the detected value is given by the sign of the real part of z.  
           [0018]    Channel coefficients can be estimated separately using standard approaches. For example, with Least Mean Squared (LMS) estimation of c 0 , one would form the time varying estimate ĉ 0 (n), where n is an index denoting symbol period, using  
             ĉ   0 ( n+ 1)= ĉ   0 ( n )+μ {circumflex over (b)} *( n )( x   0 ( n )− ĉ   0 ( n ) {circumflex over (b)} ( n ))  (5)  
           [0019]    where μ is the LMS step size. Also, {circumflex over (b)} is the detected symbol value.  
           [0020]    It can be seen that the quality of performance of a RAKE receiver is related to how well the channel delay estimator (e.g., the channel delay estimator  103 ) performs. The more accurate the estimates of signal path delays, the better the RAKE receiver will perform. An exemplary channel delay estimator  200  is illustrated in FIG. 2. The channel delay estimator  200  tests differently delayed versions of the received signal for correlation with a given spreading sequence. For each hypothesized delay, the degree of correlation determines whether the hypothesized delay represents an actual delay experienced by the received signal. To carry out this process, the exemplary channel delay estimator  200  has five “probing fingers”, each associated with one of five hypothesized delays: t 0 , t 1 , t 2 , t 3 , and t 4 . These could, for example, be equally spaced with respect to one another, such as at 0, Δt, 2Δt, 3Δt, and 4Δt, as illustrated in FIG. 2. By making Δt small, it is possible to fine tune a delay estimate and track changes in the delay. The choice of five probing fingers in this example is merely for illustration: The number of probing fingers in any particular embodiment is a design choice that can be less than, equal to, or greater than five.  
           [0021]    Except for introducing a different amount of delay, each probing finger operates in the same manner. Thus, focusing now on the probing finger associated with a delay equal to zero (i.e., no delay), the received signal is supplied to a delay unit  201  that aligns the signal to be processed in accordance with the hypothesized delay (in this case, a delay of zero). The (delayed) received signal is then passed through a matched filter  203 , which may alternatively be a bank of correlators. The matched filter  203  generates an estimate of the impulse response of the channel. This estimate is generally a complex-valued signal.  
           [0022]    The channel delay estimator  200  may operate in accordance with any of a number of standards. Merely for the purpose of illustration, and to facilitate an understanding of the invention, the exemplary channel delay estimator  200  is presumed to operate in accordance with 3GPP TS 25.211 v3.4.0 (2000-09): Third Generation Partnership Project; Technical Specification Group Radio Access Network; Physical channels and mapping of transport channels onto physical channels (FDD) (Release 1999). Of pertinence here is the fact that time multiplexing is used to define the physical channels. Section 5.3.2 of the 3GPP document defines a frame structure that comprises a number of time slots (“slots”), each associated with a different channel. During each slot, traffic channel information and control channel information are transmitted. In addition, a predetermined number of pilot bits are also transmitted during each slot. Section 5.3.3.1 of the 3GPP document specifies the format of the common pilot channel. The pilot is a predetermined sequence of bits that are known to both the transmitter and the receiver. It is this knowledge that enables the matched filter  203  to generate the estimate of the impulse response of the channel.  
           [0023]    If the channel parameters are subject to fast changes, the estimates, made for each of a number N of time slots, are summed non-coherently. This means that the absolute value of the complex signal is determined (block  205 ), and then summed with the values obtained for the signal during other time slots (summing block  207 ).  
           [0024]    Alternatively, if the channel parameters are subject to slow changes, then the channel estimates may be summed coherently, so that the absolute value block  205  would not be present. In other alternative embodiments, a combination of coherent and non-coherent averaging is also possible.  
           [0025]    The fact that the channel is fading will prevent every time slot from contributing to the estimate of the delays. However, the variations of the channel in general are such that the fading process is much faster than the changes of the delays. Thus, if we assume merely for the sake of example that, on average, there are two equally strong paths with gain h 1 , and h 2 , two peaks will be built up over time in the cumulated sum over different time slots, so long as the delays are sufficiently well separated in time.  
           [0026]    A problem exists, however, when the mutual difference in delay between multiple paths is small. In such cases, the accumulated sum can exhibit only one large peak that is situated somewhere between the true delays associated with two or more channel paths. As a consequence, only one path will be detected. This will detrimentally affect the performance of the RAKE receiver since, as mentioned above, the quality of performance of a RAKE receiver is related to how well the channel delay estimator performs.  
           [0027]    Thus, there is a need for improved methods and apparatuses for estimating the delays associated with a channel.  
         SUMMARY  
         [0028]    It should be emphasized that the terms “comprises” and “comprising”, when used in this specification, are taken to specify the presence of stated features, integers, steps or components; but the use of these terms does not preclude the presence or addition of one or more other features, integers, steps, components or groups thereof.  
           [0029]    In accordance with one aspect of the present invention, the foregoing and other objects are achieved in methods and apparatuses that estimate path delays experienced by a received signal. These estimates may, for example, be used in a RAKE receiver. Making the path delay estimates involves hypothesizing a plurality of path delays. For each of a plurality of measurement time slots and for each of the hypothesized path delays, a measurement based on the received signal is made along with a determination of whether a fade occurred. For each of the plurality of measurement time slots and for each of the hypothesized path delays, the measurement is combined with a corresponding one of a plurality of cumulative metrics only if it was determined that no fade occurred. Then, for each of the plurality of hypothesized path delays, the corresponding one of the plurality of cumulative metrics is used to determine whether the hypothesized path delay corresponds to a real path delay.  
           [0030]    In another aspect of the invention, for each of the plurality of measurement time slots, the measurement is combined with an additional cumulative metric whenever it is determined that no fades occurred for any of the hypothesized path delays. In alternative embodiments, the measurement is combined with the additional cumulative metric regardless of whether it is determined that no fades occurred for any of the hypothesized path delays. In either embodiment, the additional cumulative metric can then be used to determine whether a real path delay exists between two of the hypothesized path delays. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0031]    The objects and advantages of the invention will be understood by reading the following detailed description in conjunction with the drawings in which:  
         [0032]    [0032]FIG. 1 is a block diagram of a conventional baseband processor in a typical coherent RAKE receiver;  
         [0033]    [0033]FIG. 2 is a block diagram of an exemplary channel delay estimator;  
         [0034]    [0034]FIG. 3 is a block diagram of an exemplary delay estimator for generating gain estimates, ĥ 1  and ĥ 2  in accordance with the invention;  
         [0035]    [0035]FIGS. 4 a ,  4   b  and  4   c  are graphs that depict absolute values of exemplary received signals;  
         [0036]    [0036]FIGS. 5 a ,  5   b  and  5   c  are graphs that depict accumulated measurement sums for different fade characteristics, in accordance with an aspect of the invention;  
         [0037]    [0037]FIG. 6 is a block diagram of an alternative embodiment of a delay estimator for generating gain estimates, ĥ 1  and ĥ 2  in accordance with the invention;  
         [0038]    [0038]FIG. 7 is a graph that depicts absolute values of exemplary received signals along with hypothesized delays that are spaced relatively far apart with respect to one another; and  
         [0039]    [0039]FIG. 8 is a graph that depicts absolute values of exemplary received signals along with hypothesized delays that are spaced relatively close to one another. 
     
    
     DETAILED DESCRIPTION  
       [0040]    The various features of the invention will now be described with respect to the figures, in which like parts are identified with the same reference characters.  
         [0041]    These and other aspects of the invention will now be described in greater detail in connection with a number of exemplary embodiments. To facilitate an understanding of the invention, many aspects of the invention are described in terms of sequences of actions to be performed by elements of a computer system. It will be recognized that in each of the embodiments, the various actions could be performed by specialized circuits (e.g., discrete logic gates interconnected to perform a specialized function), by program instructions being executed by one or more processors, or by a combination of both. Moreover, the invention can additionally be considered to be embodied entirely within any form of computer readable carrier, such as solid-state memory, magnetic disk, optical disk or carrier wave (such as radio frequency, audio frequency or optical frequency carrier waves) containing an appropriate set of computer instructions that would cause a processor to carry out the techniques described herein. Thus, the various aspects of the invention may be embodied in many different forms, and all such forms are contemplated to be within the scope of the invention. For each of the various aspects of the invention, any such form of embodiment may be referred to herein as “logic configured to” perform a described action, or alternatively as “logic that” performs a described action.  
         [0042]    In accordance with one aspect of the invention, improved performance in delay estimation is achieved by making use of channel fades that affect some but not all of the delay paths. When it is detected that one or more particular paths are experiencing a fade, the estimated impulse response of the channel for that time slot is added to the accumulated values associated with only those paths that are not experiencing the fade. This and other aspects of the invention will now be described in greater detail in connection with exemplary embodiments.  
         [0043]    For simplicity, and without loss of generality, consider a two path channel, with gain h 1  and h 2 . FIG. 3 is a block diagram of an exemplary delay estimator  300  for generating gain estimates, ĥ 1  and ĥ 2  in accordance with the invention. In comparing this block diagram with that of the exemplary delay estimator  200  shown in FIG. 2, it is useful to understand that the conventional channel delay estimator  200  is depicted having a separate path for each hypothesized delay. As a consequence, the output of each absolute value block  205  is a (possibly complex) scalar. By contrast, the elements depicted in FIG. 3 are repeatedly operated, once for each hypothesized delay. As a consequence, the output of the absolute value block  303  is a complex vector, with each element in the vector representing a correlation value for a corresponding hypothesized delay. The received signal is supplied to a matched filter (or alternatively to a bank of correlators)  301  that generates an estimate of the channel impulse response, which is generally a complex-valued signal. In the exemplary embodiment, a non-coherent mean value calculation is performed, so the absolute value (block  303 ) of the channel impulse response is determined. In alternative embodiments, coherent summing of the channel impulse response estimates is performed, so that the absolute value block  303  would not be present.  
         [0044]    In accordance with an aspect of the invention, the delay estimator  300  includes a fade detector  305  that analyzes the complex signal generated by the matched filter  301  to determine, for each time slot being measured, whether any parts of the received signal associated with any of the hypothesized delays have experienced a fade. The fade detector could, for example, base its detection on channel estimates for hypothesized delays. A simple strategy could be to deem that there is no fading for a particular delay if the momentary channel estimate for this particular delay is large, while the channel is judged as being in a fading dip if the channel estimate is small. The output of the fade detector  305  is a signal that controls whether the output of the absolute value block  303  will contribute to the accumulated sum for only the first hypothesized delay  309 , to the accumulated sum for only the second hypothesized delay  311 , or to the accumulated sum that represents a delay that is in-between the first and second hypothesized delays  313 . This is depicted schematically in FIG. 3 as the output of the fade detector  305  controlling the switch position of a switch  307 . In practice, the use of a physical switch  307  is just one of a number of alternative embodiments for achieving this function. For example, in one alternative, the result of fade detection could control branching within a computer program that alternatively achieves accumulation for only the first, only the second or both hypothesized delay paths.  
         [0045]    The use of the detected fade for this purpose is based on the principle that if one path fades, a correlation peak for the other path will still be detected, enabling one to distinguish between the existence of two delay paths, and the existence of only a single delay path that is near one of the two hypothesized delay paths. Which state the switch  307  has (i.e., whether it is switch position s 1 , s 2 , or s 3 ) is preferably governed by a hypothesis of which of the paths has a sufficiently large gain. This can, for example, be given by a hypothesis scheme that roughly indicates which of the paths is/are present. For example, the possible states of the switch  307  can be given by previous channel estimates.  
         [0046]    [0046]FIGS. 4 a ,  4   b  and  4   c  depict absolute values of exemplary received signals. In FIG. 4 a , the amplitude of the signal exceeds a predetermined threshold value (below which the signal is presumed to represent only noise) only around time t 1 . Consequently, the fade detector  305  would set the switch  307  to position s 1 .  
         [0047]    In FIG. 4 b , the amplitude of the signal exceeds the predetermined threshold value only around time t 2 . Consequently, the fade detector  305  would set the switch  307  to position S 2 .  
         [0048]    In FIG. 4 c , the amplitude of the signal exceeds the predetermined threshold value around both of times t 1  and t 2 . Consequently, the fade detector  305  would set the switch  307  to position s 3 .  
         [0049]    If the above-described technique is used to measure the channel over several slots with independent fading, the various accumulated values over N slots  309 ,  311 ,  313  might look as depicted in FIGS. 5 a ,  5   b  and  5   c . More particularly, FIG. 5 a  is a graph showing the accumulated sum over N slots  309  corresponding to the switch  307  being in the s 1  position; FIG. 5 b  is a graph showing the accumulated sum over N slots  311  corresponding to the switch  307  being in the s 2  position; and FIG. 5 c  is a graph showing the accumulated sum over N slots  313  corresponding to the switch  307  being in the s 3  position. The accumulated sum information corresponding to the s 1  position of the switch  307  (e.g., FIG. 5 a ) enables the delay estimator  300  to extract the position, t 1 , of the peak value. Similarly, the accumulated sum information corresponding to the s 2  position of the switch  307  (e.g., FIG. 5 b ) enables the delay estimator  300  to extract the position, t 2 , of another peak value.  
         [0050]    In case the different multipaths were not separated enough to distinguish one from the other, the delay of the corresponding path can be found as the position of the peak of the output in FIG. 5 c  (i.e., the graph of the accumulated sum corresponding to the switch  307  being in the s 3  position).  
         [0051]    [0051]FIG. 6 is a block diagram of an alternative embodiment of a delay estimator  300 ′ for generating gain estimates, ĥ 1  and ĥ 2  in accordance with the invention. The principle of operation of this embodiment is essentially the same as that described above with respect to FIG. 3. However, in this alternative embodiment, switching is performed only between two switch positions, s 1  and s 2 . Hence, the output of the fade detector  305 ′ only has two states in the exemplary embodiment. The third summing block  313 ′ is operational at all times, and therefore need not be switched in and out.  
         [0052]    The inventive techniques described herein are most useful when two or more multipath delays are close, but not too close, in time with respect to one another. That is, if the difference between delays is relatively large, the delays are well separated from one another and the problem is rather trivial. This situation is illustrated in FIG. 7, where the distance between the hypothesized delays t 1  and t 3  is relatively large. (The hypothesized delay t 2  from previous examples is shown in FIG. 7 for the purpose of helping to illustrate the relative distance between the hypothesized delays t 1  and t 3 .) Conversely, if the delays are very close to one another, such as the distance between the hypothesized delays t 1  and t 4  depicted in FIG. 8 (with the hypothesized delay t 2  from previous examples again being illustrated for the purpose of showing the relative distance between the hypothesized delays t 1  and t 4 ), the techniques described herein will not be able to separate them; but in this case, the RAKE receiver will not require separation because the resolution of the signal is coarser than the difference between the delays. When the delays are relatively close to one another (but not too close to resolve) the techniques described herein increase the possibility for resolving the different delays by making use of the fact that, because of fading, not all delays have a strong contribution to the signal at all times.  
         [0053]    The invention has been described with reference to a particular embodiment. However, it will be readily apparent to those skilled in the art that it is possible to embody the invention in specific forms other than those of the preferred embodiment described above. This may be done without departing from the spirit of the invention.  
         [0054]    For example, in some alternative embodiments the delay resolution techniques described herein can be turned on or off based on the distance between the hypothesized delays. This function can be combined with the strategy in the fade detector.  
         [0055]    Furthermore, in order to facilitate an understanding of the invention, only the two path case has been illustrated. However, the inventive techniques described herein can easily be generalized to any number of paths having delays that are close to one another. This can be accomplished by providing a new set of fading detectors  305 , and accumulators  309 ,  311 ,  313  for each pair of multipaths.  
         [0056]    Still further, the exemplary embodiments (e.g., illustrated in FIGS. 3 and 6) include an absolute value block  303  that generates the absolute value of the complex signal supplied by the matched filter  301  in order to permit non-coherent summing to take place. Use of an absolute value block  303  is not essential to the invention, however. For example, the block  303  could alternatively generate the square of the absolute value (i.e., the power of the amplitude), or any of numerous other mathematical functions that would not significantly alter the effectiveness of the delay estimator  300 ,  300 ′. And as mentioned earlier, in still other alternative embodiments, coherent summing of the channel impulse response estimates is performed, so that the absolute value block  303  would not be present at all.  
         [0057]    Thus, the preferred embodiment is merely illustrative and should not be considered restrictive in any way. The scope of the invention is given by the appended claims, rather than the preceding description, and all variations and equivalents which fall within the range of the claims are intended to be embraced therein.