Abstract:
A quadrature modulator and demodulator which provide the requisite level of performance while minimizing power consumption. In the quadrature modulator, the I and Q signals are provided to two pairs of mixers. Each mixer in a pair of mixers modulates an I or Q signal with the respective inphase or quadrature IF sinusoid. The I and Q modulated signals from each pair of mixers are summed. The signals from the summers are provided to a third pair of mixer and modulated with the respective inphase and quadrature RF sinusoids. The signals from the third pair of mixers are summed and provided as the modulated signal. Using this quadrature modulator topology, the amplitude balance and phase error of the modulated signal are made insensitive to the amplitude imbalance and/or phase error of the quadrature splitters used to generate the IF and RF sinusoids. Furthermore, since the first two pairs of mixers and the two subsequent summers are operated at IF frequency, the performance requirements (e.g., bandwidth and linearity) of these components can be ensured while utilizing less power. The inventive concept can be further adopted for use in a quadrature demodulator.

Description:
BACKGROUND OF THE INVENTION 
     I. Field of the Invention 
     The present invention relates to communications. More particularly, the present invention relates to a novel and improved quadrature modulator and demodulator. 
     II. Description of the Related Art 
     In many modern communication systems, digital transmission is utilized because of improved efficiency and the ability to detect and correct transmission errors. Exemplary digital transmission formats include binary phase shift keying (BPSK), quaternary phase shift keying (QPSK), offset quaternary phase shift keying (OQPSK), m-ary phase shift keying (m-PSK), and quadrature amplitude modulation (QAM). Exemplary communication systems which utilize digital transmission include code division multiple access (CDMA) communication systems and high definition television (HDTV) systems. The use of CDMA techniques in a multiple access communication system is disclosed in U.S. Pat. No. 4,901,307, entitled “SPREAD SPECTRUM MULTIPLE ACCESS COMMUNICATION SYSTEM USING SATELLITE OR TERRESTRIAL REPEATERS”, and U.S. Pat. No. 5,103,459, entitled “SYSTEM AND METHOD FOR GENERATING WAVEFORMS IN A CDMA CELLULAR TELEPHONE SYSTEM”, both assigned to the assignee of the present invention and incorporated by reference herein. An exemplary HDTV system is disclosed in U.S. Pat. No. 5,452,104, U.S. Pat. No. 5,107,345, and U.S. Pat. No. 5,021,891, all three entitled “ADAPTIVE BLOCK SIZE IMAGE COMPRESSION METHOD AND SYSTEM”, and U.S. Pat. No. 5,576,767, entitled “INTERFRAME VIDEO ENCODING AND DECODING SYSTEM”, all four patents are assigned to the assignee of the present invention and incorporated by reference herein. 
     In the CDMA system, a base station communicates with one or more remote stations. The base station is typically located at a fixed location. Thus, power consumption is less important consideration in the design of the base station. The remote stations are typically consumer units which are produced in high quantity. Thus, cost and reliability are important design considerations because of the number of units produced. Furthermore, in some applications such as a CDMA mobile communication system, power consumption is critical because of the portable nature of the remote station. Tradeoffs between performance, cost, and power consumption are usually made in the design of the remote stations. 
     In digital transmission, the digitized data is used to modulated a carrier sinusoid using one of the formats listed above. The modulated waveform is further processed (e.g. filtered, amplified, and upconverted) and transmitted to the destination device. At the destination device, the transmitted RF signal is received and demodulated by a receiver. 
     A block diagram of an exemplary transmitter  100  of the prior art which is used for quadrature modulation of QPSK, OQPSK, and QAM signals is illustrated in FIG.  1 A. Transmitter  100  can be used at the base station or the remote station. Within quadrature modulator  110   a  of transmitter  100 , the inphase (I) and quadrature (Q) signals are provided to mixers  112   a  and  112   b  which modulate the signals with the inphase and quadrature intermediate frequency (IF) sinusoids, respectively. Quadrature splitter  114  receives the IF sinusoid (IF LO) and provides the inphase and quadrature IF sinusoids which are approximately equal in amplitude and 90 degrees out of phase with respect to each other. The modulated I and Q signals from mixers  112   a  and  112   b  are provided to summer  116  and combined. In many applications, the signal from summer  116  is provided to mixer  118  which upconverts the signal to the desired frequency with the radio frequency (RF) sinusoid (RF LO). Although not shown in FIG. 1A for simplicity, filtering and/or amplification can be interposed between successive stages of summers and mixers. 
     The modulated signal from mixer  118  is provided to filter  130  which filters out undesirable images and spurious signals. The filtered signal is provided to amplifier (AMP)  132  which amplifies the signal to produce the required signal amplitude. The amplified signal is routed through duplexer  134  and transmitted from antenna  136  to the destination device. 
     A block diagram of an exemplary direct quadrature modulator  110   b  is shown in FIG.  1 B. Within direct quadrature modulator  110   b,  the I and Q signals are provided to mixers  152   a  and  152   b  which modulate the signals with the inphase and quadrature RF sinusoids, respectively. Quadrature splitter  154  receives the direct RF sinusoid (direct RF LO) and provides the inphase (I LO) and quadrature (Q LO) sinusoids which are approximately equal in amplitude and 90 degrees out of phase with respect to each other. The modulated I and Q signals from mixers  152   a  and  152   b  are provided to summer  156  and combined to provide the modulated signal. 
     Quadrature modulator  110   a  performs modulation using a two steps process whereby quadrature modulation is performed at an IF frequency and upconverted to the desired RF frequency. Quadrature modulator  110   a  offers several advantages. First, quadrature splitter  114  can be more easily designed and manufactured to meet the required specification at the lower IF frequency. Second, the two sinusoids design (IF LO and RF LO) offers flexibility in the frequency plan and simplification of the filtering. 
     Direct quadrature modulator  110   b  performs the same functions as quadrature modulator  110   a . However, direct quadrature modulator  110   b  performs modulation directly at the desired RF frequency using a single step process, thereby eliminating the upconversion step. The simplicity in the design of modulator  110   b  is offset by the performance requirements of quadrature splitter  154 . In particular, it is much more difficult to design and manufacture quadrature splitter  154  having the required amplitude balance and quadrature phase at the higher RF frequency. 
     A method for generating inphase and quadrature sinusoids at RF frequency having the required performance is disclosed in U.S. Pat. No. 5,412,351, entitled “QUADRATURE LOCAL OSCILLATOR NETWORK”, and incorporated by reference herein. A block diagram of quadrature local oscillator network  170  as disclosed in U.S. Pat. No. 5,412,351 is shown in FIG.  1 C. Within quadrature local oscillator network  170 , the IF sinusoid is provided to quadrature splitter  172  which provides the inphase and quadrature IF sinusoids. The inphase IF sinusoid is provided to mixers  176   a  and  176   d  and the quadrature IF sinusoid is provided to mixers  176   b  and  176   c . Similarly, the RF sinusoid is provided to quadrature splitter  174  which provides the inphase and quadrature RF sinusoids. The inphase RF sinusoid is provided to mixers  176   b  and  176   d  and the quadrature RF sinusoid is provided to mixers  176   a  and  176   c . Mixers  176   a  and  176   b  mix the two input signals and provide the upconverted signals to summer  178   a  which combines the signals to provide the inphase direct sinusoid (I LO). Similarly, mixers  176   c  and  176   d  mix the two input signals and provide the upconverted signals to summer  178   b  which combines the signals to provide the quadrature direct sinusoid (Q LO). The inphase and quadrature direct sinusoids can be provided to mixers  152   a  and  152   b , respectively, as shown in FIG.  1 B. 
     Ideally, the inphase and quadrature sinusoids from a phase splitter are equal in amplitude and 90 degrees out of phase with respect to each other. At the RF frequency, this is difficult to achieve. For ideal quadrature splitters  172  and  174  (with no amplitude imbalance and no phase error), the inphase (I LO) and quadrature (Q LO) sinusoids are exactly equal in amplitude and 90 degree out of phase with respect to each other. Each sinusoid comprises a single tone at the difference frequency (f RF −f IF ) and no other mixing terms. The I LO and Q LO can be expressed as: 
     
       
           I   —   LO ( t )=cos(ω RF −ω IF ) t    
       
     
     
       
           Q   —   LO ( t )=sin(ω RF −ω IF ) t   (1)  
       
     
     Although quadrature local oscillator network  170  in FIG. 1C is configured to produce sinusoids at the difference frequency (f RF −f IF ), network  170  can also be reconfigured to produce sinusoids at the sum frequency (f RF +f IF ). 
     Quadrature local oscillator network  170  generates inphase and quadrature sinusoids which have improved performance over sinusoids generated by other quadrature splitters of the prior art. In particular, quadrature local oscillator network  170  substantially reduces the sensitivity of the output sinusoids to amplitude imbalance and/or phase error in quadrature splitters  172  and  174 . Amplitude imbalance and/or phase error in quadrature splitters  172  and  174  do not substantially affect the amplitude balance and quadrature phase of the output sinusoids. Instead, amplitude imbalance and phase error of quadrature splitters  172  and  174  manifest themselves as spurious signals which can be filtered. For example, an amplitude imbalance of Δ at an output of quadrature splitter  172  or  174  results in I LO and Q LO sinusoids which can be expressed as:                  I   —          LO        (   t   )         =         (     1   +     Δ   2       )          cos        (       ω   RF     -     ω   IF       )          t     +       (     Δ   2     )          cos        (       ω   RF     +     ω   IF       )          t               (   2   )                   Q   —          LO        (   t   )         =         (     1   +     Δ   2       )          sin        (       ω   RF     -     ω   IF       )          t     +       (     Δ   2     )          sin        (       ω   RF     +     ω   IF       )            t   .                                                  
     As used in this specification, an amplitude imbalance of Δ denotes that one output sinusoid from a quadrature splitter has an amplitude of 1 and the other output sinusoid has an amplitude of (1+Δ). From equation (2), each output from network  170  comprises the desired sinusoid and a spurious signal. The spurious signal has an amplitude of half the amplitude error (Δ/2) and is located at 2f IF from the desired sinusoid. This spurious signal is small in amplitude and can be filtered. More importantly, notice that the desired output sinusoids from network  170  are still amplitude balanced and in quadrature phase with each other. 
     A phase error of φ at an output of quadrature splitter  172  or  174  results in I LO and Q LO sinusoids which can be expressed as:                        I   —          LO        (   t   )         =                    cos        (       ω   RF     -     ω   IF       )            t   ·     [       1   2     +       cos        (   φ   )       2       ]         +                                  cos        (       ω   RF     +     ω   IF       )            t   ·     [       1   2     -       cos        (   φ   )       2       ]         +                                  sin        (       ω   RF     +     ω   IF       )       ·     [       sin        (   φ   )       2     ]       -                                sin        (       ω   RF     -     ω   IF       )       ·     [       sin        (   φ   )       2     ]                       Q   —          LO        (   t   )         =                    sin        (       ω   RF     -     ω   IF       )            t   ·     [       1   2     +       cos        (   φ   )       2       ]         +                                  sin        (       ω   RF     +     ω   IF       )            t   ·     [       1   2     -       cos        (   φ   )       2       ]         +                                  cos        (       ω   RF     +     ω   IF       )       ·     [       sin        (   φ   )       2     ]       +                                cos        (       ω   RF     -     ω   IF       )       ·       [       sin        (   φ   )       2     ]     .                     (   3   )                                
     As used in this specification, a phase error φ denotes that the phase of the quadrature sinusoid is (90 0 ±φ) with respect to the phase of the inphase sinusoid. From equation (3), notice that the phase error φ results in each output from network  170  comprising the desired sinusoid and two spurious signals having amplitudes of [½−cos(φ)/2] and [sin(φ)/2] and located at 2f IF  from the desired sinusoid. For small phase error φ, the spurious signals are small in amplitude. In addition, the spurious signals can be filtered since they are located at 2f IF  from the desired sinusoid. Each output from network  170  also comprises a small quadrature component of the desired sinusoid having an amplitude of sin (φ)/2. This quadrature component causes a slight rotation in the phase of the output sinusoid. However, since the inphase and quadrature output sinusoids comprise quadrature components having the sample amplitude {sin(φ)/2}, the 90 degree phase difference between the output sinusoids is maintained. 
     Although quadrature local oscillator network  170  provides the requisite performance, a major disadvantage is the power consumption. Notice in FIG. 1C that all four mixers  176  and both summers  178  operate at the RF frequency. To achieve the required circuit performance (e.g., bandwidth and linearity) at RF frequency, these circuits are biased with high current. For some applications, such as CDMA communication system, power consumption is a critical design parameter. There exists a long felt need in the industry to provide a quadrature modulator and demodulator which provide the requisite level of performance while minimizing power consumption. 
     SUMMARY OF THE INVENTION 
     The present invention is a new and improved quadrature modulator and demodulator which provide the requisite level of performance while minimizing power consumption. In the quadrature modulator of the present invention, the I and Q signals are provided to two pairs of mixers. Each mixer in a pair of mixers modulates an I or Q signal with the respective inphase or quadrature IF sinusoid. The I and Q modulated signals from each pair of mixers are summed. The signals from the summers are provided to a third pair of mixer and modulated with the respective inphase and quadrature RF sinusoid. The signals from the third pair of mixers are summed and provided as the modulated signal. 
     It is an object of the present invention to provide a quadrature modulator with improved performance. In the present invention, the inphase and quadrature IF sinusoids and the inphase and quadrature RF sinusoids are provided by two quadrature splitters. Each of the two quadrature splitters produces inphase and quadrature sinusoids which can have amplitude imbalance and/or phase error. Using the quadrature modulator topology of the present invention, the amplitude balance and quadrature phase of the modulated signal are made insensitive to the amplitude imbalance and/or phase error of the quadrature splitters. This results in improved performance of the quadrature modulator while relaxing the requirements of the quadrature splitters. 
     It is another object of the present invention to provide a quadrature modulator which utilizes the minimal amount of power while providing the requisite level of performance. In the quadrature modulator of the present invention, the first two pairs of mixers and the two subsequent summers are operated at the IF frequency. At the IF frequency, the circuit performance requirements (e.g., bandwidth and linearity) of these components can be ensured while utilizing less power. 
     It is yet another object of the present invention to provide a quadrature modulator wherein the center frequency of the modulated signal is not at the same frequency as the frequency of the IF sinusoids or the RF sinusoids. This feature reduces problems associated with sinusoidal bleedthrough from the IF or RF sinusoidals onto the modulated signal output. 
     The inventive concept of the present invention can be further adopted for use as a quadrature demodulator. In this embodiment, the RF signal is provided to two mixers which downconvert the RF signal with the inphase and quadrature RF sinusoids. The signal from each mixer is provided to a pair of mixer which demodulate the signal with the inphase and quadrature IF sinusoids. The demodulated signals from pair of corresponding mixers are provided to a summer which combines the signals to provided the demodulated I or Q baseband signal. The quadrature demodulator provides demodulated signals which are amplitude balanced and in quadrature to each other while minimizing sensitivity to the amplitude imbalance and phase error caused by the quadrature splitters used to generate the inphase and quadrature IF and RF sinusoids. Furthermore, power consumption is minimized because four of the mixers and two summers are operated at IF frequency. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The features, objects, and advantages of the present invention will become more apparent from the detailed description set forth below when taken in conjunction with the drawings in which like reference characters identify correspondingly throughout and wherein: 
     FIG. 1A is a block diagram of an exemplary transmitter of the prior art which is used for quadrature modulation of QPSK, OQPSK, and QAM signals; 
     FIG. 1B is a block diagram of an exemplary direct quadrature modulator of the prior art; 
     FIG. 1C is a block diagram of the quadrature local oscillator network of the prior art; 
     FIG. 2 is a block diagram of an exemplary quadrature modulator of the present invention which is used for quadrature modulation of QPSK, OQPSK, QAM, and various other modulation formats, including frequency modulation (FM); and 
     FIG. 3 is a block diagram of an exemplary quadrature demodulator of the present invention which is used for quadrature demodulation of QPSK, OQPSK, QAM, and various other modulation formats, including FM. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Referring to the figures, a block diagram of an exemplary quadrature modulator  210  of the present invention which is used for quadrature modulation of QPSK, OQPSK, QAM, and various other modulation formats, including FM, is illustrated in FIG.  2 . Quadrature modulator  210  can be incorporated into any transmission system, such as CDMA transmission systems. Within quadrature modulator  210 , the I signal (I IN) is provided to mixers  212   a  and  212   c  and the Q signal (Q IN) is provided to mixers  212   b  and  212   d . In the exemplary embodiment, the Q IN signal is in quadrature (90 degree phase offset) with respect to the I IN signal. Mixers  212   a  and  212   d  also receive the inphase IF sinusoid and mixers  212   b  and  212   c  also receive the quadrature IF sinusoid. Each mixer  212  modulate the respective input signal with the respective sinusoid to produce a respective I or Q modulated signal. The modulated I and Q signals from mixers  212   a  and  212   b , respectively, are provided to summer  214   a  and combined. The modulated I and Q signals from mixers  212   c  and  212   d , respectively, are provided to summer  214   b  which subtracts the Q modulated signal from the I modulated signal. The difference performed by summer  214   b  can also be achieved by summing the two inputs of summer  214   b  and inverting the Q signal to mixer  212   d  or inverting the inphase sinusoid to mixer  212   d . The signals from summers  214   a  and  214   b  are provided to mixers  216   a  and  216   b , respectively. Mixers  216   a  and  216   b  also receive the inphase and quadrature RF sinusoids, respectively, from quadrature splitter  232  and upconvert the input signals with the respective sinusoids. The upconverted signals from mixers  216   a  and  216   b  are provided to summer  218  which combines the signals to provide the modulated signal. Although not shown in FIG. 2 for simplicity, filtering and/or amplification can be provided between successive stages of mixers and summers to provide the desired signal conditioning. 
     Quadrature splitter  230  receives the IF sinusoid (IF LO) and provides the inphase and quadrature IF sinusoids which are approximately equal in amplitude and 90 degrees out of phase with respect to each other. Similarly, quadrature splitter  232  receives the RF sinusoid (RF LO) and provides the inphase and quadrature RF sinusoids which are approximately equal in amplitude and 90 degrees out of phase with respect to each other. Quadrature splitters  230  and  232  can be implemented in many embodiments. For example, the quadrature splitter can be implemented as an etched element on a circuit board using coupled transmission lines (as disclosed in the aforementioned U.S. Pat. No. 5,412,351), a Wilkinson structure, or other distributed techniques which are known in the art. The quadrature splitter can also be implemented using lump elements, such as a hybrid coupler which is commercially available. The quadrature splitter can also be implemented using a phase lock loop wherein the phase error and/or amplitude imbalance of the inphase and quadrature sinusoids are minimized by a feedback loop. In the preferred embodiment, quadrature splitters  230  and  232  are implemented using active devices. An exemplary design of a quadrature splitter using active devices is disclosed U.S. patent application Ser. No. 08/862,094, entitled “ACTIVE PHASE SPLITTER”, filed May, 22, 1997, assigned to the assignee of the present invention and incorporated by reference herein. 
     Similarly, mixers  212  and  216  can be implemented in many embodiments. The mixer can be implemented as a single balance or double balance mixer using diodes in the manner known in the art. Alternatively, the mixer can be implemented with Gilbert cell multiplier using active devices. In general, the mixer can be implemented using any non-linear device and appropriate filtering. Therefore, the various implementations of mixers  212  and  216  can be contemplated and are within the scope of the present invention. 
     Summers  214  and  218  can be implemented with passive summing elements (such as resistive networks) or active circuits (such as summing amplifiers). In the preferred embodiment, summers  214  and  218  are incorporated within mixers  212  and  216 , respectively, by proper design of mixers  212  and  216 . For example, mixers  212  and  216  can be implemented with Gilbert cell multipliers and the current outputs of a corresponding pair of multipliers are cross-connected together to provide the combined output. The implementation of a mixer pair (e.g.,  212   a  and  212   b ) and a summer (e.g.,  214   a ) using a pair of Gilbert cell multipliers is disclosed in the aforementioned U.S. Pat. No. 5,412,351. 
     For ideal quadrature splitters  230  and  232  having no amplitude imbalance and no phase error, the modulated output from quadrature modulator  210  can be expressed as:                      M        (   t   )       =                  I                   cos        (       ω   IF        t     )            cos        (       ω   RF        t     )         +     I                   sin        (       ω   IF        t     )            sin        (       ω   RF        t     )         +                                Q                   sin        (       ω   IF        t     )            cos        (       ω   RF        t     )         -     Q                   cos        (       ω   IF        t     )            sin        (       ω   RF        t     )                       =                  I                   cos        (       ω   RF     -     ω   IF       )          t     -     Q                   sin        (       ω   RF     -     ω   IF       )            t   .                       (   4   )                                
     Notice that the I and Q signals are modulated to the difference frequency (f RF −f IF ). Quadrature modulator  210  can also be configured to produce a modulated signal at the sum frequency (f RF +f IF ). This can be achieved by providing a respective inphase or quadrature sinusoid to each mixer  212  and  216  and proper combination of the I and Q modulated signals {e.g., adding the signals or taking the difference} by each summer  214  and  218 . 
     The performance of quadrature modulator  210  of the present invention can be quantify for amplitude imbalance and phase error introduced by quadrature splitters  230  and  232 . For an amplitude imbalance of Δ at an output of quadrature splitter  230  or  232 , the modulated signal can be expressed as:                      M        (   t   )       =                    I        (     1   +     Δ   2       )            cos        (       ω   RF     -     ω   IF       )          t     +       I        (     Δ   2     )            cos        (       ω   RF     +     ω   IF       )          t     +                                  Q        (     1   +     Δ   2       )            sin        (       ω   RF     -     ω   IF       )          t     ∓       Q        (     Δ   2     )            sin        (       ω   RF     +     ω   IF       )            t   .                       (   5   )                                
     From equation (5), the modulated signal comprises spurious signals having an amplitude of half the amplitude error (Δ/2) and located at 2ω IF  from the desired signal. An amplitude error of Δ in quadrature splitter  230  results in a spurious signal of −Q(Δ/2)sin(ω IF +ω IF )t in the quadrature component of the modulated signal and an amplitude error of Δ in quadrature splitter  232  results in a spurious signal of Q(Δ/2)sin(ω IF +ω IF )t. This distinction is denoted by the ∓ term in equation (5). The spurious signals are small in amplitude (Δ/2) and can be filtered since they are located at 2f IF  from the desired signal. More importantly, the desired components at the difference frequency are still amplitude balanced and in quadrature with each other. 
     A phase error of φ at an output of quadrature splitter  230  or  232  results in the modulated signal which can be expressed as:                      M        (   t   )       =                    cos        (       ω   RF     -     ω   IF       )            t   ·     [       I   2     +       I                   cos        (   φ   )         2       ]         +                                  cos        (       ω   RF     +     ω   IF       )            t   ·     [       I   2     -       I                   cos        (   φ   )         2       ]         ∓                                  sin        (       ω   RF     -     ω   IF       )            t   ·     [       I                   sin        (   φ   )         2     ]         +                                  sin        (       ω   RF     +     ω   IF       )            t   ·     [       I                   sin        (   φ   )         2     ]         +                                sin        (       ω   RF     -     ω   IF       )          t   ·     [       Q   2     +       Q                   cos        (   φ   )         2       ]         +                                  sin        (       ω   RF     +     ω   IF       )            t   ·     [       ∓     Q   2       ±       Q                   cos        (   φ   )         2       ]         ±                                  cos        (       ω   RF     -     ω   IF       )            t   ·     [       Q                   sin        (   φ   )         2     ]         ±                                cos        (       ω   RF     +     ω   IF       )            t   ·       [       Q                   sin        (   φ   )         2     ]     .                       (   6   )                                
     In equation (6), some terms are denoted by the ± designation and one term is noted by the ± designation. The upper sign of these designations denotes the sign of the term associated with a phase error φ in quadrature splitter  230  and the lower sign denotes the sign of the term associated with a phase error φ in quadrature splitter  232 . Thus, a phase error φ in either quadrature splitters  230  or  232  causes the same spurious signals. However, the sign of some spurious signals are different depending on whether the phase error φ is from quadrature splitter  230  or  232 . 
     Several observations can be made from equation (6). First, notice that the phase error φ results in the modulated signal comprising four spurious components having amplitudes of [½−cos(φ)/2] and [sin(φ)/2] and located at 2f IF  from the desired signal. For small phase error φ, these spurious signals are small in amplitude. In addition, these spurious signals can be filtered since they are located at 2f IF  from the frequency of the desired signal. The modulated signal also comprises small spurious quadrature components of the desired signals having an amplitude of sin(φ)/2. These spurious quadrature components cause a slight rotation in the phase of the I and Q components in the modulated signal. However, since these spurious quadrature components have the same amount amplitude and are in quadrature with each other, the amplitude balance and quadrature phase of the I and Q components are maintained. 
     An exemplary application of quadrature modulator  210  is for CDMA communication systems which are designed to operate at the cellular and/or personal communication service (PCS) band. In the exemplary embodiment, the IF sinusoid is generated using a phase lock loop and is fixed at nominal frequency of 130 MHz. It can be readily observed that IF sinusoids at other frequencies can be utilized and are within the scope of the present invention. In the exemplary embodiment, quadrature modulator  210  is configured to produce the modulated signal at the difference frequency (f RF −f IF ). However, quadrature modulator  210  can also be reconfigured to produce the modulated signal at the sum frequency (f RF +f IF ) and this is within the scope of the present invention. Quadrature modulator  210  can be designed to operate at the cellular band (824 MHz to 849 MHz) or the PCS band (1850 MHz to 1910 MHz). In the exemplary embodiment, the frequency of the RF sinusoid is selected to be the desired output frequency f O  plus the frequency of the IF sinusoid (f O +f IF ). This results in the modulated signal at the desired output frequency f O . 
     A block diagram of an exemplary quadrature demodulator  310  of the present invention which is used for quadrature demodulation of QPSK, OQPSK, QAM, and various other modulation formats, including FM, is illustrated in FIG.  3 . Quadrature demodulator  310  can be incorporated into any receiving system, such as those for CDMA communication systems. Within quadrature demodulator  310 , the received RF signal is provided to mixers  312   a  and  312   b . Mixers  312   a  also receives the inphase RF sinusoid, downconverts the RF signal, and provides the downconverted signal to mixers  314   a  and  314   b . Similarly, mixers  312   b  also receives the quadrature RF sinusoid, downconverts the RF signal, and provides the downconverted signal to mixers  314   c  and  314   d . Mixers  314   b  and  314   c  also receive the inphase IF sinusoid and mixers  314   a  and  314   d  also receive the quadrature IF sinusoid. Each mixer  314  demodulates the input signal with the respective IF sinusoid. The demodulated signals from mixers  314   b  and  314   d  are provided to summer  316   a  which combines the signal to provide the I output. The demodulated signals from mixers  314   a  and  314   c  are provided to summer  316   b  which subtracts the signal from mixer  314   c  from the signal from mixer  314   a  to provide the Q output. 
     Quadrature splitter  320  receives the RF sinusoid (RF LO) and provides the inphase and quadrature RF sinusoids which are approximately equal in amplitude and 90 degrees out of phase with respect to each other. Similarly, quadrature splitter  322  receives the IF sinusoid (IF LO) and provides the inphase and quadrature IF sinusoids which are approximately equal in amplitude and 90 degrees out of phase with respect to each other. Mixers  312  and  314 , summers  316 , and quadrature splitters  320  and  322  can be designed and implemented in the manner described above. As stated above, filters and/or amplifiers can be interposed between successive stages of mixers and summers to provide the desired signal conditioning. 
     The quadrature modulator and demodulator of the present invention can be implemented using many embodiments, some of which are described above. In the preferred embodiment, the quadrature modulator and demodulator are implemented within an application specific integrated circuit (ASIC) using active devices. The active devices can be bipolar-junction transistors (BJT), heterojunction-bipolar-transistor (HBT), metal-oxide-semiconductor field effect transistors (MOSFET), gallium arsenide field effect transistors (GaAsFET), P-channel devices, or other active semiconductor devices. 
     The previous description of the preferred embodiments is provided to enable any person skilled in the art to make or use the present invention. The various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without the use of the inventive faculty. Thus, the present invention is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.