Abstract:
A channelizer ( 224 ) for recovering communication channels is connected to a resampling filter ( 204 ) having an input connection ( 308 ) on which data arrives at an input data rate, and an output connection ( 310 ) on which data exits at an output data rate. The channelizer ( 224 ) further comprises an n-point discrete Fourier transform (DFT) circuit ( 208 ) coupled to the output data rate output ( 226 ) and a plurality of individual recovered channel outputs ( 210 ) coupled to the DFT circuit ( 208 ). Resampling circuitry ( 302, 306 ) in the resampling filter ( 204 ) converts the input data rate to the output data rate in a manner commensurate with a preselected communication channel spacing, communication channel output sample rate, and number of communication channels. The DFT circuit ( 208 ) may use (n-m) point overlap and the channelizer ( 224 ) may further include a polyphase filter ( 206 ) implementing n/m oversampling coupled between the resampling filter ( 204 ) and the DFT circuit ( 208 ).

Description:
BACKGROUND OF THE INVENTION  
         [0001]    Many types of wireless communication services have emerged in a relatively short period of time. Service subscribers, in turn, have quickly discovered the significant benefits in convenience and accessibility stemming from wireless communication. As a result, wireless communications services have advanced quickly into a position of popularity and profitability.  
           [0002]    Generally, a wireless communication subscriber transmits information to a base station in a “channel”. A channel represents a portion of electromagnetic spectrum having a predetermined bandwidth in which signal information resides. As one example, the European Global System Mobile (GSM) defines 200 KHz wide channels with 200 KHz spacing centered at 897.5 MHz and spanning 35 MHz of bandwidth.  
           [0003]    In certain wireless applications, a single receiver processes multiple individual channels in order to recover the signal information present in each channel. In the past, such receivers included a separate processing chain called a sub-band tuner for each channel. The sub-band tuner generally included, for example, a local IF oscillator and mixer (for converting a transmitted frequency to a first working frequency), a bandpass filter (for isolating a channel), a second IF oscillator and mixer (for downconverting the isolated channel for further processing), and an Analog to Digital converter (for digitizing the downconverted isolated channel).  
           [0004]    By processing channels individually, the receiver relaxed certain design requirements for the processing chain. For example, off the shelf low bandwidth A/D converters with 60 dB dynamic range were capable of digitizing the relatively narrow bandwidth downconverted isolated channel. However, a receiver that included multiple sub-band tuners incurred significant cost increases arising from the duplication of sub-band tuner components for each channel.  
           [0005]    As a result, designers proposed an alternative receiver implementation that used a single bulk processing chain to recover signal information from multiple channels. The bulk processing chain included an IF local oscillator and mixer (for converting a transmitted frequency to a first working frequency), a bandpass filter (for isolating multiple channels in a wide slice of bandwidth), a second IF local oscillator and mixer (for further downconverting the wide slice of bandwidth for additional processing), and a single A/D converter (for digitizing the slice of spectrum spanning the multiple channels).  
           [0006]    However, in order to recover channels from the slice of spectrum, additional processing is required. In particular, it is further necessary to isolate individual channels using a process sometimes referred to as channelization. In the past, however, there were no suitable channelizers available for cellular communication systems. As a result, the proposed bulk processing chain could not be implemented, and the more expensive and complicated individual sub-band tuner designs were adopted.  
           [0007]    A need has long existed in the industry for a cellular communications channelizer that addresses the problems noted above and others previously experienced.  
         BRIEF SUMMARY OF THE INVENTION  
         [0008]    A preferred embodiment of the present invention provides, in a cellular communications system in which terminals transmit information in cellular communication channels assigned from a predetermined spectrum, a cellular communications receiver. The cellular communications receiver includes an antenna for receiving a signal spanning multiple cellular communication channels, a downconverter coupled to the antenna for shifting the signal to an intermediate frequency, and a channelizer coupled to the downconverter for recovering the individual cellular communication channels and outputting the individual cellular communication channels on individual recovered channel outputs. The cellular communication channels may be GSM or Interim Standard cellular communication channels, as examples.  
           [0009]    The channelizer may be, for example, an n-point discrete Fourier transform (DFT) circuit, with optional m-point overlap. The variables m and n are integers with m&lt;n. In addition, the receiver may include a polyphase filter coupled to the DFT circuit, and a recovered channel output selection circuit coupled to the individual recovered channel outputs. Where the DFT circuit provides n outputs, the recovered channel output selection circuit may select a subset of recovered channel signals from the n outputs.  
           [0010]    Another preferred embodiment of the present invention provides a channelizer for recovering communication channels. The channelizer comprises a resampling filter having an input connection on which data arrives at an input data rate, and an output connection on which data exits at an output data rate. The channelizer further comprises a discrete Fourier transform (DFT) circuit coupled to the output data rate output and a plurality of individual recovered channel outputs coupled to the DFT circuit.  
           [0011]    Resampling circuitry in the resampling filter converts the input data rate to the output data rate in a manner commensurate with a preselected communication channel spacing, communication channel output sample rate, and number of communication channels. The input data rate may be, for example, a front end analog to digital converter sample rate. The channel spacing may be, as examples, GSM or Interim Standard channel spacing. The channel output sample rate may be, for example, an integer multiple of a symbol rate of a communication signal (e.g., two times the symbol rate of a GSM communication signal).  
           [0012]    The DFT circuit may be, for example, an n-point DFT circuit with m-point overlap, where m and n are integers with m&lt;n. The channelizer may further include a polyphase filter coupled between the resampling filter and the DFT circuit. The polyphase filter may implement n/m oversampling. Additionally, a recovered channel output selection circuit may be coupled to the individual recovered channel outputs for passing a subset of recovered channel signals onto subsequent processing. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0013]    [0013]FIG. 1 illustrates a cellular communications receiver.  
         [0014]    [0014]FIG. 2 shows a signal processing chain including a DFT channelizer.  
         [0015]    [0015]FIG. 3 shows a block diagram of a resampling filter.  
         [0016]    [0016]FIG. 4 shows the frequency response of a polyphase filter.  
         [0017]    [0017]FIG. 5 shows a block diagram of a polyphase filter processing block.  
         [0018]    [0018]FIG. 6 depicts an n/m oversampling polyphase filter constructed using the polyphase filter processing blocks.  
         [0019]    [0019]FIG. 7 illustrates the frequency response of a raised cosine detection filter.  
         [0020]    [0020]FIG. 8 shows a method for recovering individual communication channels from a received signal. 
     
    
     DETAILED DESCRIPTION OF THE INVENTION  
       [0021]    Turning now to FIG. 1, that figure illustrates a cellular communications channel receiver  100 . The receiver may be, for example, a cellular communication base station. The receiver  100  includes an antenna  102 , a downconverter  104 , and a channelizer  106  (described in more detail below). The channelizer  106  provides the recovered channel outputs  108 .  
         [0022]    The receiver  100  operates in a cellular communications system in which terminals transmit information in cellular communication channels assigned from a predetermined spectrum. For example, the terminals may transmit information according to the European Global System Mobile (GSM) specification, that provides 200 KHz wide channels with 200 KHz spacing centered at 897.5 MHz and spanning 35 MHz of bandwidth. The terminals may also transmit according to other types of standards, however, including the North American Interim Standard 54 or 136 standards.  
         [0023]    The downconverter  104 , in general, shifts a received signal to an intermediate frequency for further processing by the channelizer  106 . The downconverter  104  may also be accompanied by additional functionality such as filtering, automatic gain control (e.g., between 0 and 36 dB), and analog to digital (A/D) conversion. The additional functionality may include, as examples, the equalization techniques described in TRW Docket No. 12-1214, titled “Intermediate Frequency Signal Amplitude Equalizer for Multichannel Applications”, filed concurrently herewith, as Ser. No. ______, and TRW Docket No. 12-1215, titled “Amplitude Cancellation Topology for Multichannel Applications”, filed concurrently herewith, as Ser. No. ______.  
         [0024]    The downconversion process may begin, for example, with a 15 MHz wide received signal on a 897.5 MHz center subsequently 14-bit sampled at 65 Msps. The downconversion process preferably provides an intermediate frequency of one-quarter of the A/D conversion sample rate, Fs, although three-quarters and one and one-quarter are also suitable. Furthermore, in GSM systems, the intermediate frequency may be preferably offset by approximately 100 KHz.  
         [0025]    Turning next to FIG. 2, that figure illustrates a signal processing chain  200 . The processing chain includes a baseband converter  202 , a resampling filter  204 , and a polyphase filter  206 . The processing chain further includes an n-point DFT circuit  208  providing the recovered channel outputs  210 . A recovered channel output selection circuit  212  couples to the recovered channel outputs  210  and provides the selected channel outputs  214 . In one implementation of the n-point DFT circuit  208 , n=130, and the 130-point DFT is performed as a set of 13-point DFTs followed by a set of 5-point DFTs followed by a set of 2-point DFTs. Performing the DFT using such a prime number factorization generally yields an implementation that requires less resources (e.g., gates, registers, memory, and the like).  
         [0026]    Additionally, the processing chain  200  includes a raised cosine filter  216  coupled to a numeric format converter  214 . A data formatter and buffer  220  and a communication interface  222  follow. In the processing chain  200  set forth above, the channelizer  224  may be regarded as the polyphase filter  206  followed by the n-point DFT circuit  208 . The additional support circuitry  202 - 204  and  212 - 222  may be added as desired, or as required by a particular application.  
         [0027]    The baseband converter  202  provides digital mixers, oscillators, and other support circuitry that convert incoming digital samples of a received signal to complex baseband. Thus, the resultant Inphase (I) and Quadrature (Q) samples represent signal content centered around approximately DC. The received signal is preferably a wide slice (e.g., 15-35 MHz) of spectrum spanning multiple communication channels.  
         [0028]    For example, in a GSM cellular communication system, the received signal may be 15 MHz wide, thereby spanning 75 200 KHz wide channels, with 200 KHz spacing. In certain GSM implementations, only every third frequency is used. As a result, a 15 MHz portion of spectrum may in some instances include 25 active channels, rather than 75.  
         [0029]    The samples input to the baseband converter  202  are preferably 20 bit real samples. As an example, the baseband converter  202  may then downconvert by Fs/4 using a 35 tap low pass filter with approximately 0.0045 dB peak to peak pass band ripple and 91.8 dB stop band attenuation. The samples output from the baseband converter are 22 bit I and 22 bit Q samples at 65 Msps.  
         [0030]    As noted above, a front end A/D converter provides received signal samples at a predetermined A/D converter sample rate. However, the n-point DFT  208  may be designed to accommodate a different sample rate, given the communication channel spacing, communication channel output sample rate, and number of communication channels that the DFT circuit  208  recovers simultaneously. In one implementation the front end A/D converter sample rate is 65 Msps, and the preferred DFT circuit  208  sample rate is 26 Msps. As a result, the invention provides the resampling filter  204  to adapt the front end processing to the DFT processing. The front end processing and the DFT processing may thereby be designed with different goals in mind, and one need not be unduly limited by the other.  
         [0031]    Turning to FIG. 3, that figure illustrates one embodiment of a resampling filter  300 . The resampling filter  300  includes an interpolator  302 , a low pass filter  304 , and a decimator  306 . An input data rate input  308  and an output data rate output  310  are also shown. The interpolator  302  and decimator  306  function as resampling circuitry to adapt an input data rate to an output data rate.  
         [0032]    In other implementations, additional interpolation or decimation may be used, and the interpolations and decimations may be by different amounts in order to provide a suitable output data rate given an input data rate. As one example, the interpolator  302  may provide an interpolation by 2, and the decimator  306  may provide a decimation by 5. As a result, an input sample rate of 65 Msps changes to an output sample rate of (65* 2)/5=26 Msps. The low pass filter may be a 16 tap finite impulse response filter with approximately 0.021 dB peak to peak pass band ripple and 96.23 dB stop band attenuation. The samples output from the resampling filter  204  are 22 bit I and 22 bit Q samples at 26 Msps.  
         [0033]    The output of the resampling filter  300  may be served by a dual port SRAM that receives samples from the resampling filter  300  and that provides samples to the polyphase filter  206 . For the implementation described above, the dual port SRAM may be 24,992 bits in size.  
         [0034]    Referring again to FIG. 2, the output of the resampling filter  204  feeds the polyphase filter  206 . In general, a polyphase filter provides a set of filter structures in parallel, each of which may be selected at a different time to provide the different phases of the filter on incoming data. In the implementation shown in FIG. 2, the DFT circuit  208  provides n outputs, each of which is a narrowband channelized communication signal with the same bandwidth but that originated a different center frequency. The polyphase filter  206  provides the desired frequency response for each of the n communication channels. The DFT circuit  208  then replicates the frequency response across all the center frequencies at the same time.  
         [0035]    [0035]FIG. 4 presents a suitable frequency response  400  for the polyphase filter  206 . The frequency response  400  provides peak to peak pass band ripple of approximately 0.023 dB and a minimum stop band attenuation of approximately 96.7 dB using a 520 tap digital low pass filter. The samples output from the polyphase filter  206  are preferably 22 bit I and 22 bit Q samples.  
         [0036]    The polyphase filter  206  preferably implements an oversampling ratio of n/m, where n and m are integers with m&lt;n. The variable n is the number of points in the DFT circuit computation, and m is the number of new samples brought into each DFT computation. As an example, with n=130, m may be 48. In other words, the DFT circuit  208  computes a new DFT each time 48 new received signal samples arrive.  
         [0037]    Turning to FIG. 5, that figure illustrates a block diagram of a polyphase filter processing block  500 . The processing block  500  includes a delay element (e.g., a memory)  502 , a multiplier  504 , and a summer  506 . Each processing block  500  also stores a filter coefficient h(j). Signal samples enter the processing block on the sample input  508  and exit on the sample output  510 . A partial sum input  512  connects to the summer  506 , which, in turn provides the partial sum output  514 .  
         [0038]    Each signal sample stored in the delay element  502  is multiplied against the filter coefficient h(j), and added to the sample value on the partial sum input  512 . The resultant sum is output on the partial sum output  514 . When a new value is shifted into the processing block  500  on the sample input  508 , the old sample value stored in the delay element  502  propagates out on the sample output  510 .  
         [0039]    [0039]FIG. 6 depicts an oversampling polyphase filter  600  constructed using the polyphase filter processing blocks  500  (e.g., the processing blocks  602  and  604  shown labeled in FIG. 6). Each processing block includes a sample input (e.g., the sample input  606 ), a sample output (e.g., the sample output  608 ), a partial sum input (e.g., the partial sum input  610 ), and a partial sum output (e.g., the partial sum output  612 ). Processing blocks in the first column are presented with a constant value (e.g., zero) on their partial sum inputs. FIG. 6 also shows a filter input  614  and n filter outputs y(m).  
         [0040]    The polyphase filter  600  is arranged as n rows by k columns (i.e., a matrix of k*n processing blocks). Each processing block in the polyphase filter  600  stores or is otherwise associated with a filter coefficient h(j). The k*n filter coefficients h(j) implement the frequency response  400  illustrated in FIG. 4.  
         [0041]    In operation, two steps are repeated. First, m samples shift into the polyphase filter  600  (the oldest m samples are discarded) on the filter input  614 . Second, the n outputs y 0 (m) through y n−1 (m) are computed. The polyphase filter  600  thus provides an oversampling ratio of n/m with a window length of k*n and a presum ratio of k:1.  
         [0042]    The polyphase filter  206  may be served by two single port SRAMs. The polyphase filter  206  writes to one SRAM while the DFT circuit  208  reads from the other. In the implementation described above, each SRAM may be 5720 bits (i.e., large enough to hold 130 22-bit I samples and 22-bit Q samples).  
         [0043]    Referring again to FIG. 2, the DFT circuit  208  operates on the polyphase filtered signal provided by the polyphase filter  206 . In particular, the DFT circuit  208  implements an n-point DFT to provide n narrowband channelized communication signals individually on the recovered channel outputs  210 . Thus, for example, where n=130, the DFT circuit  208  provides 130 individual outputs.  
         [0044]    The DFT circuit  208  is tailored to a particular cellular communication system, communication channel spacing, communication channel output sample rate, and number of communication channels. For example, in a GSM communication system, the communication channel spacing is 200 KHz, the communication channel output sample rate may be a multiple of a symbol rate of a GSM communication signal, and the number of communication channels, N, may be 75 (assuming processing of 15 MHz of spectrum).  
         [0045]    As one example, when the communication channel spacing is 200 KHz, the communication channel output sample rate (e.g., 1625/3 Ksps) is twice the communication signal symbol rate (1625/6 Ksps), and N=75, the sample rate (Fs_in) commensurate with DFT circuit  208  operation is constrained by:  
         [0046]    N&gt;74  
         [0047]    Fs_in/N=200 KHz, and  
         [0048]    Fs_in/M=1625/3 Ksps  
         [0049]    These constraints are met by N/M=130/48, yielding a Fs_in of 26 MHz. As noted above, the resampling filter  204  adapts the front end sample rate to be commensurate with the desired DFT circuit  208  sample rate. The DFT circuit  208  preferably outputs 28 bit I and 28 bit Q output samples, for example, at 1625/3 Ksps.  
         [0050]    As noted above, the 130 point DFT may be performed first as a set of 13-point DFTs, followed by a set of 5-point DFTs, then a set of 2-point DFTs. For the implementation noted above, a 6760 bit single port SRAM may be provided to store the results of the 13-point DFT.  
         [0051]    Next, the recovered channel output selection circuit  212  determines which recovered channel signals to pass on for further processing. To that end, the selection circuit  212  may operate in response to a multibit control signal specifying the recovered channel signals to retain on the selected channel outputs  214 . The remaining recovered channel signals are discarded. The control signal may, for example, control circuitry that selectively stores in memory or discards outputs of the DFT circuit  208 , thereby selecting or retaining a subset of recovered channel signals. The output samples remain 28 bit I and 28 bit Q output samples at 1625/3 Ksps.  
         [0052]    The raised cosine filter  216  filters the subset of recovered channel signals that are retained. To that end, there may be multiple raised cosine filters  216  (e.g., one for each recovered channel signal). The raised cosine filter  216  provides, using a finite impulse response filter, a frequency response matched to the transmitter frequency response. An exemplary raised cosine filter frequency response  700  is shown in FIG. 7. The raised cosine filter  216  may be implemented, for example, using a 49 tap finite impulse response filter with approximately 0.04 dB peak to peak pass band ripple. The output samples remain 28 bit I and 28 bit Q output samples at 1625/3 Ksps.  
         [0053]    The optional numeric format converter  218  provides circuitry that converts the numeric representation used by previous processing elements to a different numeric format (if any) desired for subsequent processing. In one implementation, the numeric format converter  218  is preferably implemented using discrete logic that converts the fixed point representation used in previous processing steps to a floating point representation with a mantissa and an exponent between 2 0  and 2 15 . The output samples are preferably 12 bit I and 12 bit Q samples with 4 bits of gain information, at 1625/3 Ksps.  
         [0054]    The data formatter and buffer  220  provides circuitry to arrange the samples, after numeric formatting, into a desired output format, and to buffer the samples for transmission. For example, the data formatter and buffer  220  may prepare the samples for transmission according to a predetermined communication standard. The output samples remain 12 bit I and 12 bit Q samples with 4 bits of gain information, at 1625/3 Ksps.  
         [0055]    The communication interface  222  provides a transport mechanism for transmitting the output samples to subsequent processing. As examples, the communication interface  222  may be a serial, parallel, or network interface.  
         [0056]    In general, the processing chain  200  may be implemented in digital form in one or more ASICs. Thus, for example, where multiple recovered channel outputs  210  or selected channel outputs  214  are shown, those outputs may, in fact, be blocks of memory rather than discrete signal connections. As another example, the output selection circuit  212  may pass or block certain recovered channel signals by writing to or clearing memory areas, rather than providing outputs on discrete signal connections.  
         [0057]    Turning next to FIG. 8, that figure illustrates a method for recovering individual communication channels from a received signal. First a transmitted signal is received ( 802 ). The transmitted signal spans multiple communication channels in bandwidth. Subsequently, the processing chain  200  digitizes ( 804 ) the transmitted signal to form a digitized input signal. Next, the processing chain  200  resamples ( 806 ) the digitized received signal from an input data rate to an output data rate commensurate with a preselected communication channel spacing, communication channel output sample rate, and a number of communication channels. After polyphase filtering ( 808 ), the processing chain  200  then performs ( 810 ) an n-point DFT on the polyphase filtered resampled signal to recover individual communication channels. The processing chain may then select ( 812 ) a subset of individual recovered channels to preserve for subsequent processing.  
         [0058]    The invention thereby provides a method and apparatus for recovering many communication channels simultaneously without the need to repeat, for example, individual sub-band tuners. Thus, the invention provides significant cost and complexity benefits that grow with the number of communications channels recovered. The invention further allows a channelizer to adapt, using a resampling filter, to a wide range of input sample rates.  
         [0059]    While the invention has been described with reference to one or more preferred embodiments, those skilled in the art will understand that changes may be made and equivalents may be substituted without departing from the scope of the invention. In addition, many modifications may be made to adapt a particular step, structure, or material to the teachings of the invention without departing from its scope. Therefore, it is intended that the invention not be limited to the particular embodiment disclosed, but that the invention will include all embodiments falling within the scope of the appended claims.