Abstract:
A power-factor correction circuit for a three-phase power supply is provided. The correction circuit includes a filtering unit at the input receiving the three phases of the current, at least one inductor per phase placed downstream of the filtering unit, a rectifying bridge powering a current-chopping stage. The filtering unit includes a differential-mode filtering cell including at least one inductive circuit formed of a single magnetic material in a double E, each leg of the E being surrounded by a winding.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application is a National Stage of International Patent Application No. PCT/EP2008/068066, filed on Dec. 19, 2008, which claims priority to foreign Patent Application No. FR 07 09029, filed on Dec. 21, 2007, the disclosures of which are incorporated herein by reference in their entirety. 
     FIELD OF THE INVENTION 
     The present invention relates to a power-factor correction circuit for a three-phase power supply. The invention applies notably to the field of power electronics, in particular the production of three-phase power supply units. 
     BACKGROUND OF THE INVENTION 
     A power electronic circuit sometimes introduces a reactive power causing a phase shift of the power supply current and voltage. Moreover, one or more conversion stages present in said electronic circuit, for example a stage for rectifying the voltage, cause deformations of the input current, consequently degrading the power factor. The power electronic circuit therefore requires, at its input, the addition of a correction circuit in order to increase the power factor. 
     On the one hand, the correction circuit, indicated by the acronym PFC in the rest of the description with reference to the expression “Power Factor Correction”, must put the current and the voltage back into phase. On the other hand, the PFC circuit must limit the harmonic distortions of the input current. Therefore, a PFC circuit must at least meet two constraints simultaneously: obtain a high power factor and a good quality of the induced-current harmonic distortion. The new standards, notably concerning purity in current shape are increasingly strict, as shown, for example, by the chapters relating to conducted emissions of the MIL-STD-461E standard of the American Defense Department. 
     To meet the aforementioned strict constraints with applications powered by a three-phase current, it is natural to juxtapose three PFC stages, one for each phase of the electric current. However, although this solution makes it possible to achieve good performance, both in terms of harmonic distortion and in terms of power factor, it culminates in a complex architecture, notably because of the balancing difficulties between the three PFC stages. Moreover, the resultant circuit is bulky because of the gearing-down of the components to be used. 
     An alternative solution using the principle of the PFC circuits of the “boost” type is shown in  FIG. 1 . It is a conventional correction circuit  100  for a three-phase power supply comprising a first filtering unit  101  dedicated to the low frequencies, a hexaphase rectifying bridge  102 , a second filtering unit  103  dedicated to the high frequencies, and a voltage step-up stage  104 , which comprises an inductor  105 , a controlled switch  106 , and a freewheel diode  107  powering a reservoir capacitor C. The value of the inductor  105  is chosen to be sufficiently large for the circuit  100  to operate in continuous mode. The capacitor C is a reserve of energy making it possible to power a user circuit, modeled in  FIG. 1  by a load  110 . This conventional correction circuit  100  makes it possible, without having recourse to three PFC stages, to significantly increase the power factor of the circuit. However, since the value of the inductor  105  has to be high to obtain an acceptable smoothing of the current, the inductive component chosen to fulfill this role is often very bulky. Moreover, in practice, the architecture of this circuit shows its limits in quality of the harmonic distortions; it does not make it possible to satisfy the requirements of the strictest standards. 
     Other solutions have been proposed, notably a circuit shown in the patent referenced U.S. Pat. No. 6,984,964 by the applicant Delta Electronics Inc. This circuit, designed for a three-phase power supply, makes it possible to obtain low levels of harmonic distortion while maintaining a high power factor. However, this circuit is particularly costly, because it requires the use of a Digital Signal Processor or DSP, and a complex programmable circuit or CPLD (“Complex Programmable Logic Device”), in order to control the backflows of current toward the input of the circuit notably when the neutral of the three-phase network is not connected to the circuit. Moreover, it is necessary to have 3 distinct PFC functions, one per phase in order to perform the “PFC” function making it possible to obtain all at the same time a power factor close to the unit combined with a low input-current harmonic distortion, for example in order to satisfy the requirement of the CE101 test of the MIL-STD-461E standard. 
     SUMMARY OF THE INVENTION 
     One object of the invention is to propose a PFC circuit making it possible to comply with the strict requirements relating to the power factor and to levels of harmonic distortions, while limiting the size of said PFC circuit and its cost. Accordingly, the subject of the invention is a correction circuit of the power factor of a circuit for a three-phase electric network comprising a filtering unit at the input receiving the three phases of the current, at least one inductor per phase placed downstream of the filtering unit, a rectifying bridge powering a current-chopping stage, characterized in that the filtering unit comprises a differential-mode filtering cell comprising at least one inductive circuit formed of a single magnetic material in a double E, each leg of the E being surrounded by a winding, the values of the inductors being chosen so that said correction circuit operates at the boundary between the continuous mode and the discontinuous mode. 
     According to one embodiment, the filtering unit comprises a first common-mode filtering cell, associated in series with a first differential-mode filtering cell and a second differential-mode filtering cell, the filtering unit also comprising a second common-mode filtering cell comprising a mid-point output and inserted between the first differential-mode filtering cell and the second differential-mode filtering cell. 
     According to one embodiment, the second differential-mode filtering cell comprises simple inductors and the assembly of the differential-mode inductors is incorporated into a molded resin block, said assembly consisting of the inductive modules and of the simple inductors of the first differential-mode filtering cell and of the simple inductors of the second differential-mode filtering cell. 
     A further subject of the invention is a power supply unit comprising a power-factor correction circuit as described above. 
     Unlike the PFC circuits of the prior art, the circuit according to the invention is unitary, that is to say that it comprises a single PFC circuit operating in three-phase mode. This feature has several advantages. On the one hand, it makes it possible to reduce the overall size of the circuit by a factor of 3. On the other hand, it makes it possible to solve the problems of balancing the phases, whether or not the neutral of the three-phase network is connected to the circuit. Moreover, an additional architecture requiring computing modules such as a DSP or a CPLD is no longer necessary. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Other features will appear on reading the following detailed description given as a non-limiting example with respect to the appended drawings which represent: 
         FIG. 1 , a correction circuit architecture according to the prior art, the figure has already been explained, 
         FIG. 2 , the architecture of an embodiment of a correction circuit optimized according to the invention, 
         FIG. 3   a , the illustration of the first phase of the current-chopping cycle carried out by a correction circuit according to the invention, 
         FIG. 3   b , the illustration of the second phase of the current-chopping cycle carried out by a correction circuit according to the invention, 
         FIG. 4   a , the shape of the current I LS  passing through an inductor of the correction circuit according to the invention, and an illustration of the shape of the phase current I PH  obtained with said circuit, after an optimization of the inductor value, 
         FIG. 4   b , a closer view of the inductor current I LS  shown in  FIG. 4   a,    
         FIG. 4   c , an illustration of the shape of the phase current I PH  obtained with a correction circuit according to the invention, 
         FIG. 5 , a schematic diagram of a filtering unit contained in a correction circuit according to the invention, 
         FIG. 6   a , a top view of a filtering unit contained in a correction circuit according to the invention, 
         FIG. 6   b , two sections, in side view, of a filtering unit contained in a correction circuit according to the invention, 
         FIG. 7   a , a top view of an inductive module present in the filtering module of a correction circuit according to the invention, 
         FIG. 7   b , a side view of an inductive module present in the filtering module of a correction circuit according to the invention. 
     
    
    
     DETAILED DESCRIPTION 
     For the purposes of clarity of the description, the same reference numbers in the various figures designate the same elements. 
       FIG. 2  shows the architecture of an embodiment of the correction circuit according to the invention. 
     The correction circuit  200  of  FIG. 2  comprises a filtering unit  201  comprising three inputs  201   a ,  201   b ,  201   c  and three outputs  201   d ,  201   e ,  201   f . The filtering unit  201  is powered by three current phases, a first phase V R  on the first input  201   a , a second phase V S  on the second input  201   b , and a third phase V T  on the third input  201   c . Moreover, the filtering unit  201  is connected to ground  201   t . This filtering unit  201  will be explained in detail below in  FIGS. 3 ,  5 ,  6   a  and  6   b.    
     A filtering stage  211 , making it possible to reject high-frequency components generated by the chopping of the current carried out by the PFC stage  214  described below, is placed in series at the output of the filtering unit  201 . More precisely, in the example of  FIG. 2 , the first output  201   d  of the filtering unit  201  is connected to its second output  201   e  via a first capacitor C RS , its second output  201   e  is connected to its third output  201   f  via a second capacitor C ST , and its third output  201   f  is connected to its first output  201   d  via a third capacitor C RT . According to another embodiment, the high-frequency filtering stage  211  is incorporated into the filtering unit  201 . 
     An inductive stage  212  is placed at the output of the high-frequency signal filtering stage  211 . More precisely, in the example of  FIG. 2 , the first output  201   e  of the filtering unit  201  is connected to the first terminal  202   a  of a first inductor L R ; the second output  201   e  of the filtering unit  201  is connected to the first terminal  203   a  of a second inductor L S ; the third output  201   f  of the filtering unit  201  is connected to the first terminal  204   a  of a third inductor L T . 
     The inductors L R , L S  and L T  are connected to a diode bridge  213  for rectifying the voltage. More precisely, in the example of  FIG. 2 , the second terminal  202   b  of the first inductor L R  is connected to the anode of a first diode D 1 ; the second terminal  203   b  of the second inductor L S  is connected to the anode of a second diode D 2 ; the second terminal  204   b  of the third inductor L T  is connected to the anode of a third diode D 3 . Moreover, the anode of the first diode is connected to the cathode of a fourth diode D 4 ; the anode of the second diode D 2  is connected to the cathode of a fifth diode D 5 ; the anode of the third diode is connected to the cathode of a sixth diode D 6 . The diode bridge  213  comprises two outputs  213   a ,  213   b . The cathodes of the first diode D 1 , of the second diode D 2  and of the third diode D 3  are connected together at the first output  213   a  of the diode bridge  213 . The anodes of the fourth diode D 4 , of the fifth diode D 5  and of the sixth diode D 6  are connected together at the second output  213   b  of the diode bridge  213 . 
     The first output  213   a  of the diode bridge  213  is connected to the second output  213   b  via a controlled switch  214 . In the example, the controlled switch  214  is formed by an MOSFET (Metal Oxide Semiconductor Field Effect Transistor) transistor  205  the drain  205   a  of which is connected to the first output  213   a  of the diode bridge  213 , the source  205   b  of which is connected to the first terminal  206   a  of a shunt resistor  206 , and the gate  205   c  of which is controlled by a chopping signal generator  207 . In the example, this generator  207  is a pulse width modulation generator, a signal at the low state causing a disabling of the MOSFET  205 , a signal at the high state controlling the flow of the current through the MOSFET  205 . The frequency of transmission of said pulses is very much higher than the frequency of the current originating from each of the input phases V R , V S  and V T . For example, the frequency of transmission of the pulses is of the order of 125 kHz for a network current at 400 Hz. The second terminal  206   b  of the shunt resistor  206  is connected to the second output  213   b  of the diode bridge  213 . This resistor  206  of very low value makes it possible to measure the current originating from the source  205   b  of the MOSFET  205  in order to adapt the width of the pulses transmitted by the generator  207 . 
     Moreover, a freewheel diode D RL  is placed so that its anode is connected to the first output  213   a  of the diode bridge  213  and to the drain  205   a  of the transistor  205 . 
     Finally, a reservoir capacitor Cs is placed at the end of the circuit in order to store the energy necessary for the user circuit  110  to be powered, including in the event of a transient cut-out of the input AC network; the first terminal of this capacitor Cs being connected to the cathode of the freewheel diode D RL , the second terminal of this capacitor being connected to the second output  213   b  of the diode bridge  213 . 
     The correction circuit of  FIG. 2  operates according to a two-phase cycle: a first phase during which the controlled switch  214  is closed, and a second phase during which the controlled switch  214  is open. 
     The first phase is illustrated in  FIG. 3   a . The controlled switch  214  being closed, a short circuit is formed at the branch comprised between the first output  213   a  of the diode bridge  213  and its second output  213   b . The current, shown in  FIG. 3   a  by an arrow Fi, therefore flows through this branch  213   a ,  213   b  as a short circuit and allows the inductors L R , L S , L T  to store magnetic energy. In parallel, the freewheel diode D RL  is disabled and the current passing through said diode D RL  is zero. 
     This magnetic energy stored by the inductors L R , L S  and L T  is restored by the capacitor Cs during the second phase of the cycle, as illustrated in  FIG. 3   b . The current, represented in the figure by an arrow Fi, is transmitted in full to the freewheel diode D RL . 
     The values of the inductors L R , L S  and L T  are optimized in order to limit the need for low-frequency signal filtering carried out by the filtering unit  201 , without degrading the input-current harmonic distortion. 
     The values of the inductors L R , L S , L T  are preferably chosen so that the correction circuit  200  operates at the boundary between the continuous mode and the discontinuous mode, in other words, the inductors L R , L S , L T  finish discharging at the moment of beginning the first phase of the cycle. The lower the value of each of the inductors L R , L S , L T , the more attenuated must be the inversion of the high-frequency current originating from the chopping carried out by the controlled switch  214  in order to maintain the high-frequency conducted emission requirements and hence the greater must be the attenuation of the filtering carried out by the filtering unit  201  in order to limit the harmonic distortions of the input current. 
     By contrast, the higher the value of the inductors L R , L S  and L T , the higher the risk of degrading the low-frequency harmonic distortion (that is to say the frequency of the network and its near harmonics). If the values for the inductors L R , L S  and L T  are too high, they would lead to a degradation of the envelope signal of the input current. Then, in order to preserve the shape of this envelope signal, it would be necessary to significantly increase the low-frequency filtering volume. 
     Thus, the optimized value of the inductors L R , L S  and L T  is chosen to reconcile the following two requirements: to obtain a low line-current distortion at the frequencies close to the frequency of the network and to satisfy the high-frequency requirements with respect to the pollution generated by the chopping. 
       FIG. 4   a  illustrates, over time t, the shape of the phase current I PH  of the second phase of the network and the shape of the current I LS  passing through the second inductor L S , the shape of the current passing through the other inductors L R  and L T  being similar. The current I PH  is represented with respect to a first axis  401 , while the current of inductor I LS  is represented with respect to a second axis  402 . When the current sine curve is close to the maximum, a plateau  403 , forming a continuous component of the current I LS , appears. The obtained shape of the inductor current I LS , in particular the plateau  403  and the phase relative to the current I PH  is due to the abovementioned optimization of the value of the inductors L R , L S , L T . With this optimization, the current I PH  is marked by a harmonic distortion equal to approximately 15%, or a significant improvement over a circuit of the prior art (from 40% to 25% distortion, depending on the power to be delivered), as illustrated in  FIG. 4   a.    
       FIG. 4   b  is a closer view of the shape of the current I LS  of inductor L S  previously shown in  FIG. 4   a . The current increases linearly during the first phase  411  of the chopping cycle. Then, during the second phase  412 , the inductor discharges and the current reduces as far as to cancel itself out, the current again increasing after the elapsing of the period T of the cycle. The current in each inductor L R , L S , L T  therefore takes the shape of a succession of triangles. 
     As shown below, the distortion of the current is further diminished by improvements made to the filtering unit  201  placed at the input of the correction circuit according to the invention. The shape of the current benefiting from these improvements is shown in  FIG. 4   c.    
     The current I LS  remains unchanged (relative to the reading in  FIG. 4   a ) but the shape of the line current is again optimized so that the resultant harmonic distortion is of the order of 5%, namely of a kind to satisfy the strictest requirements, notably those defined by the test marked CE101 of the MIL-STD-461 E standard of the American Defense Department. 
     With respect to the architecture shown in  FIG. 1 , the architecture of  FIG. 2  has the advantage of better eliminating the undesirable low-frequency harmonic rays, particularly those of the fifth harmonic and of the seventh harmonic. On the other hand, since the current is modulated at a high rate (for example 70% to 100%) at the chopping frequency, the high-rank harmonic rays are greater than for a correction circuit operating in continuous mode. Therefore, relative to the circuit shown in  FIG. 1 , the low-frequency filtering can be lightened, while the high-frequency filtering must be improved. Nevertheless, carrying out high-frequency filtering is much less of a disadvantage in terms of space occupancy than low-frequency filtering which requires bulky components. 
       FIG. 5  shows a block diagram of a filtering unit  201  placed at the input of the correction circuit of the embodiment of  FIG. 2 . The filtering unit  201  comprises a first common-mode filtering cell  501 , which is associated in series with a first differential-mode filtering cell  502 , which is associated in series with a second common-mode filtering cell  503 , which is associated in series with a second differential-mode filtering cell  504 . 
     The first common-mode filtering cell  501  comprises a conventional filtering module  510  place at the head of the filtering unit  201  as close as possible to the input connections  201   a ,  201   b ,  201   c . This module  510  comprises three inputs  510   a ,  510   b ,  510   c  and three outputs  510   d ,  510   e ,  510   f , the first input  510   a  of said module  510  being connected to the first input  201   a  of the filtering unit  201 , the second input  510   b  of said module  510  being connected to the second input  201   b  of the filtering unit  201 , the third input  510   c  of said module  510  being connected to the third input  201   c  of the filtering unit  201 . Each of the outputs, respectively  510   d ,  510   e ,  510   f , of the filtering module  510  is connected to ground via a capacitor, respectively C 1 , C 2 , C 3 . 
     The first differential-mode filtering cell  502  makes it possible to filter the low-frequency components of the current, that is to say in a frequency band extending substantially from 30 Hz to 15 kHz. This cell  502  comprises an original structure making it possible to reduce its volume compared with a conventional filtering cell. Specifically, said cell  502  comprises one or more three-phase inductive modules  511 ,  511 ′,  511 ″, in the example, three, associated in series in order to obtain a sufficiently high inductor value. Said inductive modules  511 ,  511 ′,  511 ″ comprise three inputs  511   a ,  511   b ,  511   c  and three outputs  511   d ,  511   e ,  511   f  each of the inputs corresponding to a current phase of a three-phase network, each of the outputs also. The inductive modules  511 ,  511 ′,  511 ″ are shown in detail in  FIG. 7 . Moreover, the first differential-mode filtering cell  502  also comprises, for each current phase, a simple inductor  512 ,  512 ′,  512 ″, each of these inductors being associated in series with the three-phase inductive module(s)  511 ,  511 ′,  511 ″. In the example, a first inductor  512  is connected to the first output  511   a ″ of the third inductive module  511 ″, the second inductor  512 ′ is connected to the second output  511   b ″ of the third inductive module  511 ″ and the third inductor  512 ″ is connected to the third output  511   c ″ of the third inductive module  511 ″. The association of simple inductors  512 ,  512 ′,  512 ″ with the three-phase inductive modules  511 ,  511 ′,  511 ″ makes it possible to efficiently complete the filtering carried out by said modules  511 ,  511 ′,  511 ″, notably for the highest frequencies of the low-frequency template, in the example, in the range from 10 to 15 kHz. Specifically, the value of the inductive modules  511 ,  511 ′,  511 ″ begins to reduce toward 10 kHz. A supplement to the low-frequency filtering is therefore carried out by placing the aforementioned simple inductors  512 ,  512 ′,  512 ″ in series. These simple inductors  512 ,  512 ′,  512 ″ are achieved by windings of the same nature as the windings  514 ,  514 ′,  514 ″ dedicated to the filtering of the high-frequency components of the second differential-mode filtering cell  504 , the inductor value of these windings beginning to reduce for much higher frequencies. 
     Therefore, the simple inductors  512 ,  512 ′,  512 ″ take over from the inductive modules  511 ,  511 ′,  511 ″ for the highest frequencies of the low-frequency template involved in the requirements of the standards relating to conducted emissions, notably the CE101 test of the MIL-STD-461E standard. 
     Moreover, capacitors C 4 , C 5 , C 6  are placed at the output of the simple inductors  512 ,  512 ′,  512 ″ so that a first capacitor C 4  connects the output of the first simple inductor  512  with the output of the second simple inductor  512 ′, a second capacitor C 5  connects the output of the second simple inductor  512 ′ with the output of the third simple inductor  512 ″ and a third capacitor C 6  connects the output of the first simple inductor  512  with the output of the third simple inductor  512 ″. These capacitors make it possible to promote the rejection of the frequency components generated by the chopping of the current. 
     The second differential-mode filtering cell  504  makes it possible to eliminate the interference frequency components originating from the chopping of current as illustrated in  FIGS. 3   a ,  3   b , that is to say the components the frequency of which is equal to or greater than the chopping frequency. 
     Moreover, compared with a conventional filtering unit, the filtering unit  201  used in the correction circuit according to the invention comprises a second common-mode filtering cell  503  inserted between the first  502  and the second  504  differential-mode filtering cell. This second filtering cell  503  acts as an isolation buffer between the two differential-mode filtering cells  502 ,  504 . Mid-point outputs  513   a ,  513   b ,  513   c  for each of the three phases of the current are produced. These mid-point outputs connect common-mode capacitors C 7 , C 8 , C 9  of the second common-mode filtering cell  503  to ground. 
     These capacitors C 7 , C 8 , C 9  are necessary for keeping to the requirements of the input standards of common mode type but they must in no circumstances interfere with the operation of the chopping stage  214  situated downstream of the filtering unit  201 . The mid-point outputs  513   a ,  513   b ,  513   c  of the second common-mode filtering cell  503  are therefore added so that the winding portion situated downstream of the capacitors C 1 , C 2 , C 3  serves as an shock inductor so as to prevent the high-frequency components of the current, components originating from the chopping by the controlled switch  214 , from re-closing via the common-mode capacitors C 1 , C 2 , C 3  of the second common-mode filtering cell  503 . In the absence of this filtering cell  503 , current-flow interference loops may appear between said capacitors and the current chopping stage  205 ,  206 . A mid-point output outlet  513   a ,  513   b ,  513   c  is therefore produced for each phase winding in order to allow said capacitors to be wired to ground. 
       FIG. 6   a  shows a top view of an embodiment of a filtering unit  201  placed at the input of a correction circuit according to the invention. The inputs and outputs of the inductors  512 ,  512 ′,  512 ″,  514 ,  514 ′,  514 ″ and of the inductive modules  511 ,  511 ′,  511 ″ are shown by dots  601 , that is all the differential-mode windings. In the example, the differential-mode filtering cells  502 ,  504  are incorporated into a molded block  602 . Measurements mentioning the dimensions in mm are given in  FIG. 6   a  as an indication in order to give a better idea of the size of the filtering unit  201 . 
     By virtue, notably, of the original structure of the first differential-mode filtering cell  502 , the size of the filtering unit  201  is small. As an example, the filtering unit  201 , dimensioned for a user circuit  110  ( FIG. 2 ) requiring a power of 1 kW, has the following dimensions: a length of less than 300 mm, a width equal to 55 mm, and a height equal to 48 mm. 
       FIG. 6   b  shows two sections  600 ,  600 ′ of the filtering unit  201  seen from the side. A first section  600  shows the filtering unit  201  at the inductive modules  511 ,  511 ′,  511 ″ and a second section shows the filtering unit  201  at the common-mode inductors  510 ,  513 . 
     In the example of  FIGS. 6   a  and  6   b , the assembly of the differential-mode inductors is incorporated into the molded resin block  602 , these inductors being the inductive modules  511 ,  511 ′,  511 ″, the simple inductors  512 ,  512 ′,  512 ″ of the first differential-mode filtering cell  502  and the simple inductors  514 ,  514 ′,  514 ″ of the second differential-mode filtering cell  504 . 
     On the other hand, the common-mode inductors  510 ,  513  being made of a ferrite material having to be protected from the mechanical stresses, these two common-mode inductors  510  and  513  are mounted in a second stage by bonding to the molded resin block  602 . Therefore, the filtering unit  201  is formed by the association of the molded block  602  with the common-mode inductors  510  and  513 . To hold the assembly of the molded block  602  to the structure of an equipment chassis, a first series of five struts  603  is produced. A printed circuit (not shown in the figures and on which the common-mode and differential-mode capacitors are notably installed) is electrically connected to the molded block  602 . Also, a second series of three struts  604  holds said printed circuit mechanically to the molded block  602 . 
     In the embodiment shown in  FIGS. 6   a  and  6   b , the assembly of the filtering unit  201  is shielded by a casing, in the example made of μ-metal. The assembly thus shielded is suitable for satisfying the low-frequency radiating requirements, notably for the H magnetic field. 
       FIGS. 7   a  and  7   b  give details of the structure of an inductive module  511  used in the first differential-mode filtering cell  502 , the other inductive modules  511 ′,  511 ″ being similar.  FIG. 7   a  shows a top view of the module  511  while  FIG. 7   b  shows a side view of the same module  511 . Three elementary windings are wound around one and the same magnetic core. The inductive module  511  is a three-phase inductor: each of the elementary windings is wound around each leg of a double-E circuit  705 ,  705 ′, the two E-shaped circuits being assembled so that the legs of each “E” are placed one facing the other in order to form a semblance of an “8”. In the example, the gap between the two circuits  705 ,  705 ′ in the shape of an “E” is equal to 0.3 mm. After mounting of the magnetic circuit, one or more turns of tape  706  is positioned around the three elementary windings. 
     As an indication, in the example, each inductor formed by the elementary winding  701 ,  702 ,  703  is 1.35 mH for a current of 3.5 effective amperes. For one and the same inductor value, the inductive module  511  therefore forms a three-phase winding of smaller size that three separate windings. 
     The electric definition (the three-phase nature of the winding) of the inductive module  511  in the filtering unit  201  makes it possible to further reduce the distortion of the input current. The inclusion of the inductive module  511  makes it possible to go from 15% of harmonic current distortion, that is the line current I pH  obtained by virtue of the optimization of the value of the inductors L R , L S , L T , to a harmonic current distortion of, in the example, between 8% and 5%, as illustrated in  FIG. 4   c.    
     An advantage of the PFC circuit according to the invention is that it has only one current chopping control, thus simplifying its architecture and its operation. Moreover, the PFC circuit according to the invention is of small size when compared with the three-phase PFC circuits of the prior art, which, for onboard equipment, is sometimes a decisive advantage. 
     As an illustration, for a primary network producing a three-phase current of 115 VAC at 400 Hz, the correction circuit:
         makes it possible, for an input power of 700 W, to obtain a power factor equal to 0.99, and a main SHD (Single Harmonic Distortion) harmonic of less than 9% for the H5 rays of the fifth harmonic and H7 rays of the seventh harmonic (that is to say the H5 and H7 rays respectively at 2000 Hz and 2800 Hz);   makes it possible, for an input power of 1 kW, to obtain a power factor equal to 0.95, an SHD equal to 6% for the ray of the fifth harmonic and 2% for the ray of the seventh harmonic, the distortion ratio for the other rays being less than 2%.       

     These performances are notably compatible with complying with the CE101 test of the MIL-STD-461E standard.