Abstract:
Disclosed is a very low frequency test generator for generating a high voltage having a low frequency in order to test the insulation of capacitive loads, in particular power cables. Said VLF test generator comprises an oscillator part which generates a high voltage that has a high frequency and is modulated with a lower frequency at an output, and a demodulator which is connected to the oscillator part, demodulates the high voltage, and recovers the low frequency therefrom. A discharge resistor for the capacitive load is connected in parallel to the demodulator, said discharge resistor conducting back to the aforementioned output.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application is a National Phase application of International Application No. PCT/AT2009/000211 filed May 20, 2009 which claims priority to Austrian Patent Application No. A 856/2008 filed May 28, 2008. 
     BACKGROUND 
     The present invention relates to a VLF test generator for generating a high voltage with a low frequency for testing the insulation of capacitive loads, in particular power cables, having an oscillator part which generates at an output a high voltage which has a high frequency and is modulated with a low frequency, and a demodulator connected thereto for demodulating the high voltage and recovering the low frequency therefrom. 
     Testing with high voltages of a very low frequency (VLF) in the tenth of a hertz range is now established practice for power testing of the insulation of highly capacitive loads such as buried cable systems. Unlike mains frequency or higher frequency test voltages, VLF test voltages bring about only slight reactive power in the capacitive load, such that the test generator may be made correspondingly smaller; and, in comparison with previously used direct voltage tests, VLF test voltages prevent any build-up of harmful space charges and residual charges in the cable system, which on subsequent operation could result in dielectric breakdowns. 
     Generating suitable VLF test voltages in the high voltage range, i.e. of up to several hundred kilovolts, is however not in any way straightforward, as high voltage transformers are not feasible for such low frequencies. The most varied circuits have thus already been proposed for VLF test generators, but all of them either involve highly complex circuitry or have costly or fault-prone components. 
     DE 103 33 241 B, for example, discloses a VLF test generator of the initially stated type which uses a variable transformer with a motor-driven adjustment means in order to amplitude-modulate a mains frequency high voltage by periodic adjustment of the transformer. The amplitude-modulated high voltage is stepped up and then, with the assistance of a demodulator, the modulation frequency is recovered as a VLF high voltage. The demodulator is formed by a diode rectifier whose conducting-state direction can be changed over and which changes over at each half-wave of the low frequency in order to reverse the capacitive load with each half-wave. A switchable discharging resistor is connected in parallel to the capacitive load in order to assist the reversal. Such a load-parallel discharging resistor does however result in elevated power loss and/or requires additional switching electronics with correspondingly increased costs, weight and cooling requirements. 
     The object of the invention is to overcome the disadvantages of the known prior art and to provide a VLF test generator for the generation of low-frequency high voltages which can be produced simply and inexpensively, has a low weight for mobile, on-site use, has a low power loss and accordingly a low cooling capacity. 
     SUMMARY 
     This object is achieved with a VLF test generator of the above-stated kind which is distinguished according to the invention in that a discharging resistor for the capacitive load which leads back to the stated output is connected in parallel to the demodulator. 
     In comparison with conventional circuits with a permanent load-parallel discharging resistor, the circuit according to the invention has a substantially lower power loss; and in comparison with solutions with a switchable load-parallel discharging resistor, the solution according to the invention does not require a separate switch, because the discharging resistor related to the output potential of the resonant circuit always comes particularly strongly into effect when the interference product in the resonant circuit has its beat node and thus approaches zero potential. As a consequence, it is possible to make significant savings in costs, weight, power loss and cooling requirements. 
     Demodulation of the low frequency may be effected with any demodulator circuit known in the art. One solution which is particularly simple in circuit design terms is achieved, as is known per se from the cited document DE 103 33 241 B, if the demodulator co-uses the capacitive load and reverses the latter by means of a rectifier in step with the low frequency. In this case, a particularly advantageous embodiment of the invention involves simply connecting the discharging resistor in parallel to the rectifier. 
     Any rectifier circuit known in the art may also be used for the rectifier. It is particularly advantageous if, as is known per se from DE 103 33 241 B, the rectifier comprises two antiparallel diode branches provided with switches, the switches alternately changing over between the diode branches. In this case, the discharging resistor may simply be connected in parallel to the two diode branches, which is a solution with minimal component requirements. 
     In a further preferred embodiment of the invention, each of the stated diode branches is formed by a chain of diodes and interposed semiconductor switches, an individual resistor being connected in parallel to each diode and each semiconductor switch, said individual resistors all jointly forming the stated discharging resistor. In this way, the number of components required may be still further reduced and an elevated electric strength achieved. 
     A further preferred feature of the invention provides that, on changeover, the switches are briefly closed simultaneously and overlappingly. In this way, the transient response of the generator output voltage on changeover of the rectifier can be minimised. 
     It is furthermore particularly advantageous if, according to a further feature of the invention, a control device is additionally provided for the oscillator part, which control device reduces the amplitude of the high voltage at the end of each second quarter of the period of the low frequency, in order to assist discharge of the capacitive load via the discharging resistor. 
     Any oscillator circuit known in the art may also be used for the oscillator part, which oscillator circuit is capable of generating a low-frequency modulated high voltage, for example electric motor driven variable transformers, as are known per se from DE 103 33 241 B. It is, however, particularly favourable if the oscillator part comprises two oscillators, the oscillator frequencies of which differ from one another by twice the stated low frequency, and a resonant circuit supplied interferingly by the oscillators which is tuned to the oscillator frequencies for voltage superelevation of the interfering oscillator frequencies, the demodulator outcoupling the low frequency generated by the interference from the resonant circuit and applying it to the load. At variance with all known solutions, this embodiment is based on the new approach of making use of the interference or beat between two slightly differing oscillators for generating a low-frequency modulation in a resonant circuit, which approach simultaneously brings about a voltage rise of the interference product. In this manner, an output voltage of very high amplitude and very low frequency can be generated with surprisingly few components. Fault-prone mechanical elements or complex power electronics with elevated cooling requirements are entirely unnecessary. Due to its low weight, space requirement and robustness, the VLF test generator according to the invention is particularly suitable for mobile use in on-site insulation testing, for example of buried power cables. 
     It is particularly favourable if, according to a further feature of the invention, the oscillators supply the resonant circuit via at least one transformer, whereby electrical isolation of the oscillators from the resonant and thus high voltage circuit may be achieved and negative repercussions on the oscillators of transient phenomena on the high voltage side may be prevented; furthermore, an inductive load for the oscillator outputs may be provided in this manner, as is required, for example, by the stated power modules. 
     One particularly advantageous embodiment of the invention is distinguished in that the transformers are simultaneously used to step up the oscillator output voltages for supply to the resonant circuit. A further increase in the output voltage of the VLF test generator may be achieved in this manner. 
     The voltage rise in the resonant circuit in particular also makes it possible to use conventional controllable semiconductor inverters for the oscillators, as are known in the form of “power modules” and are capable, for example, of generating any desired output voltage profiles of up to 400 V from a mains frequency supply voltage of 400 V; solely as a result of the voltage rise in the resonant circuit, it is possible to obtain therefrom output voltages in the range of a few tens of kV. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The invention will be explained below with reference to an exemplary embodiment shown in the drawings, in which: 
         FIG. 1  is a circuit diagram of a first embodiment of a VLF test generator; 
         FIG. 2  shows the interference product, the resonance voltage U R , which occurs in the resonant circuit; 
         FIG. 3  shows the low-frequency modulation of resonance voltage U R , magnified not-to-scale; 
         FIG. 4  is a circuit diagram of a second embodiment of a VLF test generator with the discharge circuit according to the invention; 
         FIG. 5  shows downregulation of the oscillator output voltages in every second quarter period of the low frequency and switching profiles of the rectifier from  FIG. 4  to assist load discharge; 
         FIGS. 6 and 7  show two alternative embodiments for the diode branches of the rectifier and the discharging resistor of  FIG. 4 ; and 
         FIGS. 8 to 10  show alternative circuit variants for connecting the oscillators to the resonant circuit. 
     
    
    
     DETAILED DESCRIPTION 
       FIG. 1  shows a test generator  1  which generates a high voltage U S  in the range from several tens to several hundred kV and with a very low frequency (VLF) in the range from a few hertz and below. The VLF test generator  1  serves to test the insulation of a capacitive load  2 , for example an underground high voltage cable. Such loads  2  generally have a capacitance in the range of up to a few μF. The further measuring arrangement for testing the insulation of the load  2  after application of the low-frequency high voltage U S , in particular for measuring the output voltage, for accompanying diagnostic measurements, such as dissipation factor measurements or partial discharge measurements etc., is not of relevance here and is not shown. 
     The test generator  1  is substantially composed of an oscillator part  3  and a demodulator  4  connected thereto. The oscillator part  3  generates at an output  5  a high voltage U R  which has a relatively high frequency and is amplitude-modulated with the stated low frequency, and the demodulator  4  demodulates the modulation product U R  in order to obtain therefrom a low-frequency high voltage U S  as the generator output voltage and to apply it to the load  2 . 
     As shown in  FIG. 1 , the oscillator part  3  comprises two oscillators  6 ,  7 , the oscillator frequencies f 1 , f 2  of which differ by twice the desired low frequency f S  of the generator output voltage U S , i.e. f 2 −f 1 =2f S . The oscillator frequencies f 1 , f 2  are preferably substantially higher than the conventional mains frequency of electrical power distribution networks (50 or 60 Hz), specifically in general in the range from 100 Hz to 10 kHz, preferably in the range from 500 Hz to 5 kHz, and particularly preferably around 1 kHz, for example f S =1000.0 Hz and f 2 =1000.2 Hz. 
     The oscillators  6 ,  7  jointly supply in series connection a resonant circuit formed by a choke  8  and a capacitor  9 . Due to the mutual superposition or interference of the oscillator output voltages U 1 , U 2 , an interference product is established in the resonant circuit  8 ,  9  which may be regarded an oscillation of frequency 
                 f   R     =         f   1     +     f   2       2       ,         
hereafter designated the resonance voltage U R , which is amplitude-modulated with a low-frequency beat U S  of frequency
 
                 f   S     =         f   2     -     f   1       2       ,         
as shown in  FIGS. 2 and 3 .
 
     Since, in the stated example, f S =0.1 Hz, the 10,000 fold higher frequency resonance voltage U R  is only visible in  FIG. 2  as an area; for greater clarity, the resonance voltage U R  is shown in  FIG. 3  with a not-to-scale magnified period. 
     Due to the series connection of the oscillators  6 ,  7 , the amplitude of the excitation voltage U 1 +U 2  of the resonant circuit  8 ,  9  is twice the amplitude of the individual oscillator output voltages U 1 , U 2 . The resonant circuit  8 ,  9  is tuned to the frequency f R  of the excitation voltage U 1 +U 2 , such that the resonance voltage U R  at the resonant circuit is raised by the quality Q of the resonant circuit relative to the exciting oscillator voltages U 1 +U 2  and thus reaches 2·Q times one of the oscillator output voltages U 1 , U 2 . 
     The quality Q of the resonant circuit  8 ,  9  is preferably between 10 and 100, particularly preferably between 50 and 80. In this manner, due to the voltage interference and voltage rise in the event of resonance of the resonant circuit  8 ,  9 , it is possible to generate a resonance voltage U R  in the range from for example 60-80 kV from oscillator output voltages U 1 , U 2  in the range from 3-400 V. 
     In order to outcouple the low-frequency high voltage U S  from the resonant circuit  8 ,  9 , the demodulator  4  shown here co-uses the load  2 , specifically by reversing the latter via a switched rectifier  10 - 13  in step with the low frequency f S . The demodulator  4  comprises for this purpose two antiparallel diode branches  10 ,  11 , which, at each half-wave of the low frequency f S , are alternately connected to the output  5  of the resonant circuit  8 ,  9  by means of corresponding switches  12 ,  13 . 
     In order to prevent any jump in voltage as the generator output voltage U S  passes through zero, this being brought about for example by voltage drops in the rectifier  10 - 13  and/or residual charges in the load  2 , according to the prior art a discharging resistor  14  may be connected in parallel to the load  2 . The discharging resistor  14  may be connected in parallel to the load  2  permanently or, with the assistance of a switch (not shown), only during the phase in which the output voltage U S  is passing through zero. 
     Instead of such a (switchable) load-parallel discharging resistor  14  according to the prior art, which entails elevated power loss and/or additional switching electronics, the following discharge circuit according to  FIG. 4  is used. 
       FIG. 4  shows an alternative embodiment of the VLF generator of  FIG. 1 , identical reference numerals denoting identical parts. As an alternative to  FIG. 1 , in this embodiment the two oscillators  6 ,  7  are connected in parallel to one another and interfere via their output currents, but the series connection of  FIG. 1  may also be used. 
     In the embodiment of  FIG. 4 , a discharging resistor  15  is arranged in parallel to the demodulator  4  (or more precisely its switchable diode branches  10 ,  11 ) and discharges the load  2  towards the potential of the output  5  of the oscillator part  3 . As a result, the discharging resistor  15  is particularly effective precisely during the phase when the output voltage U S  is passing through zero, because at that point the output  5  is also tending towards zero due to the nodes of the resonant frequency U R . 
     According to  FIG. 5 , the effectiveness of the discharging resistor  15  may be increased in that, in the in each case second quarters b, d of the four quarters a-d of the period of the low frequency f S , the output voltages U 1 , U 2  of the oscillators  6 ,  7  are slightly reduced, specifically in particular in the final part b′, d′ of the quarters b, d, such that the envelope curve of the resonance voltage U R  no longer has an exactly sinusoidal profile at that point. The actual time profile of this voltage reduction is here controlled with the assistance of a controller  16  which measures the generator output voltage U S  in a feedback control circuit such that, taking account of the voltage drop in the diode branches  10 ,  11  and switches  12 ,  13  and the residual charges in the load  2 , overall a maximally sinusoidal profile of the generator output voltage U S  is obtained. 
     The discharging resistor  15  connected in parallel to the demodulator  4  results in a degree of crosstalk of the high frequency f R  to the output frequency f S . Appropriate dimensioning of the discharging resistor  15  and control of voltage reduction in zones b′, d′ can minimise this effect to such an extent that the degree of distortion or harmonic distortion of the generator output voltage U S  is for example below 5% THD. 
     Discharge of the load  2  on changeover of the rectifier  10 - 13  may be further assisted by another measure.  FIG. 5  shows the time profile of the switching schematics S 12 , S 13  of the switch  12 ,  13 . As may be seen, actuations S 12 , S 13  of the switches  12 ,  13  during changeover may overlap slightly, specifically such that the switches  12 ,  13  are both simultaneously closed for a brief period (zones a′, c′) immediately after the resonance voltage U R  has passed through zero. As a result, the transient response of the generator output voltage U S  may be minimised on changeover of the rectifier  10 - 13  and thus a still better approximation to an ideal sinusoidal profile may be achieved. 
     In the example shown of a VLF period 1/f S  lasting  10  s, the closure overlap a′, c′ preferably amounts to approx. 0.1 s. In general, the closure overlap a′, c′ is in the range from a few thousandths to a few hundredths of 1/f S . 
       FIG. 6  shows a first practical embodiment of the diode branch  10  and the discharging resistor  15  (diode branch  11  is a mirror image). As is known in high voltage engineering, the diode branch  10  is preferably formed by a chain of individual diodes  10 ′,  10 ″ etc. and interposed individual semiconductor switches  12 ′,  12 ″ etc. Each diode  10 ′,  10 ″ is interconnected with a serial current-limiting resistor  17 ′,  17 ″ etc., a parallel testing resistor  18 ′,  18 ″ etc. and a parallel protection capacitor  19 ′,  19 ″ etc. 
     Actuation of the semiconductor switches  12 ′,  12 ″ is schematically symbolised by a control line  21  provided with resistors  20 ′,  20 ″ etc., via which line the switching signal S 12  is supplied; the actual actuation circuits for the semiconductor switches  12 ′,  12 ″ are known to a person skilled in the art and are not shown in greater detail here. In order to increase electric strength, the discharging resistor  15  is made up of series-connected individual resistors  15 ′,  15 ″ etc. 
     As shown in  FIG. 7 , the parallel testing resistors  18 ′,  18 ″ of the diodes  10 ′,  10 ″ may be co-used to form the discharging resistor  15 . For this purpose, resistors  22 ′,  22 ″ etc. are connected in parallel to the semiconductor switches  12 ′,  12 ″, which resistors, together with the testing resistors  18 ′,  18 ″ and the resistor chain located parallel thereto of the mirror image diode branch  11  (not shown), form the discharging resistor  15 . 
       FIGS. 8 to 10  show various practical embodiments of the oscillators  6 ,  7  and their connection to the resonant circuit  8 ,  9  (shown only in part). The oscillators  6 ,  7  are here in each case formed by semiconductor inverters, the output voltages of which may be adjusted by microprocessor control to any desired frequency or amplitude (“power modules”). 
     In the embodiment of  FIG. 8 , each oscillator  6 ,  7  supplies the resonant circuit  8 ,  9  via a dedicated high voltage transformer  23 ,  24 . The transformers  23 ,  24  serve various purposes: for electrical isolation of the oscillators  6 ,  7  from the resonant circuit  8 ,  9 ; for electrical isolation from one another, for providing an inductive load for the oscillators  6 ,  7 ; and for additional stepping up of the oscillator output voltages U 1 , U 2  for excitation of the resonant circuits  8 ,  9 . For example, oscillator output voltages U 1 , U 2  of approx. 400 V may be stepped up therewith to a resonant circuit excitation voltage of approx. 4 kV, such that a generator output voltage U S  of approx. 400 kV may be achieved with a resonant circuit of quality Q=100. 
     In the embodiment of  FIG. 9 , the two oscillators  6 ,  7  share a common transformer  25 , each oscillator supplying a dedicated primary winding  25 ′,  25 ″ of the transformer  25 , which are arranged in series on the transformer core, such that here too voltage interference of the oscillator output voltages U 1 , U 2  is obtained. 
       FIG. 10  finally shows a further embodiment in which a single high voltage transformer  26  is used, to the primary winding of which are connected the oscillators  6 ,  7  in galvanic parallel connection (or series connection, not shown), such that interference of the oscillators here proceeds by current interference (or voltage interference, not shown) in the primary circuit. 
     As symbolised by the arrow  27  in  FIG. 10 , the oscillator frequency f 2  of the one oscillator  7  may be derived from the oscillator frequency f 1  of the other oscillator  6 , whereby elevated constancy of the beat frequency f S =(f 2 −f 1 )/2 may be achieved. Alternatively, the oscillator frequencies f 1 , f 2  of the oscillators  6 ,  7  may also be derived from a common clock generator  28 , see  FIG. 9 . 
     If no particularly high output voltage U S  is required, the voltage-transforming high voltage transformers  23 - 26  may also be replaced by matching transformers for impedance matching and electrical isolation. 
     The invention is not limited to the embodiments shown but instead encompasses all variants and modifications, in particular any desired combinations of the exemplary embodiments shown, which fall within the scope of the appended claims,