Abstract:
The present invention relates to a ferroelectric memory device comprising a cell array block, a data bus unit and a control circuit unit. The cell array block has a bitline structure including a main bitline and a plurlaity of sub bitlines. The main bitline is connected to a column selection controller, and the plurality of sub-bitlines have both terminals connected to the main bitline, respectively, and connected to a plurality of unit cells. The data bus unit is connected to the column selection controller. The control circuit unit includes a sense amplifier array connected between a data I/O buffer and a sense amplifier data bus connected to the data bus unit. A plurality of the cell array blocks are arranged like a matrix. The control circuit unit is disposed in a first center line of symmetry wherein the first center line is parallel to the main bitline, and the data bus unit is disposed in a second center line of symmetry wherein the second center line is vertical to the main bitline. 
     The layout of the disclosed memory device allows capacitance load of a data bus to be minimized in a highly integrated circuit, thereby embodying a high-speed FRAM.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to a ferroelectric memory device, and more specifically, to an effective arrangement of cell arrays and control circuits to improve the integration of a chip and minimize delay factors of signals. 
     2. Description of the Related Art 
     Generally, a ferroelectric random access memory (hereinafter, referred to as ‘FRAM’) has attracted considerable attention as next generation memory device because it has a data processing speed as fast as a DRAM (Dynamic Random Access Memory) and conserves data even after the power is turned off. 
     The FRAM includes capacitors similar to the DRAM, but the capacitors have a ferroelectric substance for utilizing the characteristic of a high residual polarization of the ferroelectric substance in which data is not low even after eliminating an electric field applied thereto. 
     FIG. 1 is a characteristic curve illustrating a hysteresis loop of a general ferroelectric substance. As shown in FIG. 1, a polarization induced by an electric field does not vanish but keeps some strength (‘d’ or ‘a’ state) even after the electric field is cleared due to existence of a residual (or spontaneous) polarization. These ‘d’ and ‘a’ states may be assigned to binary values of ‘1’and ‘0’ for use as a memory cell. 
     FIG. 2 is a structural diagram illustrating a unit cell of the FRAM device. As shown in FIG. 2, the unit cell of the conventional FRAM is provided with a bitline BL arranged in one direction and a wordline WL arranged in another direction vertical to the bitline BL. A plateline PL is arranged parallel to the wordline and spaced at a predetermined interval. The unit cell is also provided with a transistor T 1  having a gate connected to an adjacent wordline WL and a source connected to an adjacent bitline BL, and a ferroelectric capacitor FC 1  having the first terminal of the two terminals connected to the drain terminal of the transistor T 1  and the second terminal of the two terminals connected to the plateline PL. 
     FIG. 3 a  is a timing diagram illustrating a write mode of the conventional FRAM. 
     Referring to FIG. 3 a , when a chip enable signal CSBpad applied externally transits from a high to low level and simultaneously a write enable signal WEBpad also transits from a high to low level, the array is enabled to start a write mode. Thereafter, when an address is decoded in a write mode, a pulse applied to a corresponding wordline transits from a “low” to “high” level, thereby selecting the cell. 
     In order to write a binary logic value “1” in the selected cell, a “high” signal is applied to a bitline BL while a “low” signal is applied to a plateline PL. In order to write a binary logic value “0” in the cell, a “low” signal is applied to a bitline BL while a “high” signal is applied to a plateline PL. 
     FIG. 3 b  is a timing diagram illustrating a read mode of the conventional FRAM. Referring to FIG. 3 b , when a chip enable signal CSBpad externally transits from a “high” to “low” level, all bitlines are equalized to a “low” level by an equalization signal before selection of a required wordline. 
     After each bitline is deactivated, an address is decoded to transit a signal on the required wordline from a “low” to “high” level, thereby selecting a corresponding unit cell. A “high” signal is applied to a plateline of the selected cell to cancel a data Qs corresponding to the logic value “1” stored in the FRAM. If the logic value “0” is stored in the FRAM, a corresponding data Qns will not be destroyed. 
     The destroyed and non-destroyed data output different values, respectively, according to the above-described hysteresis loop characteristics. As a result, a sense amplifier senses logic values “1” or “0”. In other words, as shown in the hysteresis loop of FIG. 1, the state moves from ‘d’ to ‘f’ when the data is destroyed while the state moves from ‘a’ to ‘f’ when the data is not destroyed. 
     As a result, the destroyed data amplified by the enabled sense amplifier outputs a logic value “1” while the non-destroyed data amplified by the sense amplifier outputs a logic value “0”. The original data is destroyed after the sense amplifier amplifies the data. Accordingly, when a “high” signal is applied to the required wordline, the plateline is disabled from “high” to “low”, thereby recovering the original data. 
     SUMMARY OF THE INVENTION 
     Cell arrays and control circuits should be effectively arranged to embody a highly integrated FRAM operating at a high speed. 
     Accordingly, it is a first object of the present invention to maximize the efficiency of a layout by arranging adjacent circuits such as a pad array, a sense amplifier array and an address buffer in a center of symmetry of a cell array block, and symmetrically arranging data bus units perpendicular to the other center of symmetry of the cell array block. 
     It is a second object of the present invention to allow data to be effectively restored and written in the FRAM by controlling a sense amplifier using a column selection signal. 
     It is a third object of the present invention to supply a VPP to each cell array block at a high speed by dividing VPP-related circuits involved in the cell operation into a gate control-related VPP circuit of a small capacity and a VPP pump circuit of a large capacity, and effectively arranging them. 
     It is a fourth object of the present invention to provide a layout of a connection portion between the data bus unit and the cell array block, which increases process margin and signal transmission efficiency and minimizes a required area. 
     There is provided a ferroelectric memory device, comprises a cell array block, a data bus unit and a control circuit unit. The cell array block has a bitline structure including a main bitline and a plurlaity of sub bitlines. The main bitline is connected to a column selection controller, and the plurality of sub-bitlines have both terminals connected to the main bitline, respectively, and connected to a plurality of unit cells. The data bus unit is connected to the column selection controller. The control circuit unit includes a sense amplifier array connected between a data I/O buffer and a sense amplifier data bus connected to the data bus unit. A plurality of the cell array blocks are arranged like a matrix. The control circuit unit is disposed in a first center line of symmetry wherein the first center line is parallel to the main bitline, and the data bus unit is disposed in a second center line of symmetry wherein the second center line is vertical to the main bitline. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a characteristic curve illustrating a hysteresis loop of a general ferroelectric substance. 
     FIG. 2 is a structural diagram illustrating a conventional FRAM cell device. 
     FIGS. 3 a  and  3   b  are timing diagrams illustrating read and write operations of a conventional FRAM. 
     FIG. 4 is a block diagram of a FRAM according to the present invention. 
     FIG. 5 is a structural diagram illustrating a control circuit unit and a cell array block of FIG.  4 . 
     FIG. 6 is a circuit diagram illustrating a sense amplifier array unit and a sense amplifier data bus unit included in the control circuit unit of FIG.  4 . 
     FIG. 7 is a diagram illustrating a first example of connection between the sense amplifier array unit and a data bus unit of FIG.  6 . 
     FIGS. 8 a  and  8   b  are diagrams illustrating a second example of connection between the sense amplifier array unit and the data bus unit of FIG.  6 . 
     FIGS. 9 a  and  9   b  are diagrams illustrating a third example of connection between the sense amplifier array unit and the data bus unit of FIG.  6 . 
     FIG. 10 is a block diagram illustrating a global controller and a local controller for controlling sense amplifiers. 
     FIGS. 11 a  and  11   b  are circuit diagrams illustrating the sense amplifier of FIG.  10 . 
     FIGS. 12 and 13 are timing diagrams of the sense amplifier of FIGS. 11 a  and  11   b.    
     FIG. 14 is a structural diagram illustrating a cell array block of FIG.  4 . 
     FIG. 15 is a circuit diagram illustrating a main bitline pull-up controller of FIG.  14 . 
     FIG. 16 is a circuit diagram illustrating a column selection controller of FIG.  14 . 
     FIG. 17 is a circuit diagram illustrating a main bitline load controller and a sub cell block of FIG.  14 . 
     FIGS. 18 a  and  18   b  are timing diagrams illustrating read and write operations of the sub cell block of FIG.  17 . 
     FIG. 19 is a layout of the connection portion between a data bus unit and a column selection controller according to the present invention. 
     FIG. 20 is a block diagram illustrating a VPP supply circuit for supplying a VPP to a cell array block of the FRAM according to the present invention. 
     FIG. 21 is a structural diagram illustrating a VPP driving circuit of FIG.  20 . 
     FIG. 22 is a timing diagram illustrating the operation of the VPP driving circuit of FIG.  21 . 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     The present invention will be described in detail with reference to the accompanying drawings. 
     FIG. 4 is a block diagram of a FRAM according to the present invention. Four cell array blocks  300  are arranged as a 2×2 matrix format in a chip. A control circuit unit  100  including an address input pad, a buffer, a decoder and a sense amplifier array is arranged between a first row and a second row of the matrix. A data bus unit  200  is arranged between a first column and a second column of the matrix. A bitline (not shown) of a cell array block  300  is connected to a data bus line in the data bus unit  200  through a column selection controller  310 . 
     FIG. 5 is a detailed structural diagram illustrating the control circuit unit of FIG.  4 . The control circuit unit  100  comprises a sense amplifier array  100 , a column address pad  122 , a column address buffer  121 , a column address decoder  120 , a row address pad  112 , a row address buffer  131 , a row address decoder  130 , an I/O pad  141 , a data I/O buffer  140 , and a chip controller  150 . The sense amplifier array  110  includes a plurality of sense amplifiers. A column address is inputted into the column address pad  122 , stored in the column address buffer  121 , and decoded by the column address decoder  120 . A row address is inputted into the row address pad  112 , stored in the row address buffer  131 , and decoded by the row address decoder  130 . Data is inputted or outputted at the I/O pad  141 , and stored in the data I/O buffer  140 . The chip controller  150  controls the operation of a chip. 
     An output signal from the row address decoder  130  controls a wordline/plateline driver  320  to output a driving voltage to a wordline and a plateline in read/write operations. 
     FIG. 6 is a block diagram illustrating the sense amplifier array  110  in the control circuit unit  100  of FIG. 4, and a sense amplifier data bus unit  160  for connecting a data bus unit  200  to the sense amplifier array  110 . Each sense amplifier in the sense amplifier array  110  shares the data bus unit  200 . 
     There are various methods for connecting the data bus unit  200  to the sense amplifier array  110 , which are explained below. 
     FIG. 7 is a diagram illustrating a first example of connection between the sense amplifier array unit  110  and the data bus unit  200  of FIG.  6 . Each sense amplifier  111  in the sense amplifier array  110  is connected to a sense amplifier data bus line  161  in the sense amplifier data bus  160 . The sense amplifier data bus line  161  is directly connected to a data bus line  210  in the data bus unit  200 . 
     FIG. 8 a  is a diagram illustrating a second example of connection between the sense amplifier array unit  110  and the data bus unit  200  of FIG.  6 . Switches SW 1  and SW 2  are arranged apart in the middle portion of the data bus unit  200 . The sense amplifier data bus line  161  corresponding to each sense amplifier  111  is directly connected to the corresponding data bus line  210  between the switches SW 1  and SW 2 . The sense amplifier  11  is connected to the right or left side of the data bus unit by the complementary switching operations of SW 1  and SW 2 . 
     FIG. 8 b  shows the enlarged connection portion of FIG. 8 a.    
     FIG. 9 a  is a diagram illustrating a third example of connection system between the sense amplifier array unit  110  and the data bus unit  200  of FIG.  6 . Unlike the second example, the data bus unit  200  is divided into a first data bus unit on the left side and a second data bus unit on the right side in the third example. The sense amplifier data bus is also divided into a first sense amplifier data bus connected to the first data bus unit and a second sense amplifier data bus connected to the second data but unit. 
     A data bus line  162  in the first sense amplifier data bus is directly connected to a first data bus line  210 -L in the first data bus unit. A data bus line  163  in the second sense amplifier data bus is directly connected to a second data bus line  210 -R in the second data bus unit. 
     FIG. 9 b  shows the enlarged connection portion of FIG. 9 a . A first switch SW 1  is connected to each data bus line  162  in the first sense amplifier data bus. A second switch SW 2  is connected to each data bus line  163  in the second sense amplifier data bus. The switches SW 1  and SW 2  are connected to the same port of the sense amplifier  111 . The structure of the sense amplifier  11  will be explained in detail later. The sense amplifier  11  is connected to the first sense amplifier data bus line  162  or the second sense amplifier data bus line  163  by the complementary switching operations of SW 1  and SW 2 . 
     FIG. 10 is a block diagram illustrating a global controller and a local controller for controlling the sense amplifier according to the present invention. A control signal inputted to the sense amplifier  11  is generated from a global controller  170  and a local controller  180 . The global controller  170  outputs a common control signal into all sense amplifiers  111  in the sense amplifier array. The local controller  180 , which is located in each sub sense amplifier array of a sense amplifier array, outputs a common control signal into a plurality of the sense amplifiers  111  in the sub sense amplifier array. The local controller  180  is controlled by a column address bit Yi&lt;n&gt; while the global controller  170  generates a control signal regardless of a column address bit. 
     The sense amplifiers  111  selected by the column address bit start read or write modes. Since a read mode is necessarily accompanied with a restore mode, the sense amplifier  111  selected by the column address bit performs a restore or write operation. However, the rest sense amplifiers  111 , which are not selected by the column address bit, start only a read mode with a restore mode. 
     As a result, the global controller  170  outputs a signal which is commonly used in read and write modes. The local controller  180  generates a control signal for performing read and write modes into the selected sense amplifier  111 , and a control signal for performing only a read mode into the unselected sense amplifier  111 . 
     The specific operation of the sense amplifier  111  and its control signals will be explained below. 
     FIG. 11 a  is a circuit diagram illustrating the sense amplifier  111  of FIG.  10 . The sense amplifier  111  comprises a data line pull-up controller  400 , an amplification unit  500 , and an I/O controller  600 . 
     The data line pull-up controller  400  pulls up a voltage of a data line to a VCC in response to a control signal DBPU_C. The data line is connected to a sense amplifier data bus line. 
     The amplification unit  500  comprises a first comparator  510 , a second comparator  530 , an equalizer  520 , and a storage unit  540 . The first comparator  510  compares a signal of a data line with that of a reference line, and outputs a high level signal when the signal of the data line is higher than that of the reference line. The second comparator  530  outputs an opposite level signal to the first comparator  510 . The equalizer  520  equalizes a voltage from an output unit of the first comparator  510  with that of the second comparator  530 . The storage unit  540  includes two input terminals connected through the first comparator  510  and the second comparator  530 , and each switch  550  and  560 . 
     The I/O controller  600  includes a first path  610 , a second path  620 , a third path  630  and a fourth path  640 . The first path  610  transmits data inputted from a data I/O buffer (not shown) into the storage unit  540 . The second path  620  outputs data stored in the storage unit  540 . The third path  630  transmits an output signal from the second path  620  into the data I/O buffer. The fourth path  640  transmits the output signal from the second path  620  into the data line. 
     The storage unit  540  stores output signals from the first comparator  510  and the second comparator  530  in a read mode, thereby performing a restore operation after the read operation. In a write mode, the storage unit  540  stores data transmitted from the first path  610 , and transmit the data into data lines of the second path  620  and the fourth path  640 , thereby allowing data to be written in the memory cell. Here, the restore operation is similarly performed to the write operation. 
     FIG. 11 b  is a circuit diagram illustrating another example of the sense amplifier  111  of FIG.  10 . The major function of the example shown in FIG. 11 b  is the same as that of the sense amplifier shown in FIG. 11 a . However, a PMOS transistor  521  is used herein instead of the equalizer  520  of FIG. 11 a . The PMOS transistor  521  has a gate to receive a control signal identical with the control signal of the data line pull-up controller  400 , a source connected to the VCC, and a drain connected to an output terminal of the first comparator  510 . 
     FIGS. 12 and 13 are timing diagrams of the sense amplifier of FIGS. 11 a  and  11   b . FIG. 12 is a timing diagram illustrating a write mode when the column address bit Yi&lt;n&gt; is activated. FIG. 13 is a timing diagram illustrating a write mode when the column address bit Yi&lt;n&gt; is inactivated. 
     Referring to FIG. 12, if a write enable signal WEB is activated, a WSN becomes “high” and the first path  610  of FIG. 11 a  is activated. A WHSN becomes “low” and the second path  620  of FIG. 11 a  is inactivated (t 0 ). Thereafter, SEN 1 , STGN and SEN 2  are activated, and a signal of the data line is stored in the storage unit  540  of FIG. 11 a  (t 2 ). If the column address bit Yi&lt;n&gt; is activated, the SEN 2  and the switches  550  and  560  of FIG. 11 a  are inactivated. Next, data inputted in the I/O buffer is stored in the storage unit  540  of FIG. 11 a  (t 3 ). If the WHSN becomes “high” and the second path is activated (t 5 ), data stored in the storage unit  540  of FIG. 11 a  is outputted to the data line through the activated (t 4 ) fourth path  640  of FIG. 11 a.    
     Referring to FIG. 13, although the write enable signal WEB is activated, the WSN is maintained at a low level, and the first path  610  is inactivated. The WHSN is maintained at a high level, and the second path is activated. If the SEN 1 , the SEN 2  and the STGN are activated, the value of the data line is read, and stored in the storage unit  540  (t 2 ). Next, A LSN is activated, and the fourth path is activated (t 4 ). Then, the value stored in the storage unit is outputted into the data line. That is, when the column address bit Yi&lt;n&gt; is not activated, the restore operation is only performed. 
     As described above, the global controller  170  generates a signal that is identically operated when the column address bit is selected and unselected. The local controller  180  generates that is not identically operated. Referring to FIGS. 12 and 13, the control signals SEN 1 , SEN 2 , LSN, LSP, STGN, STGP, SEQN and SEQP are generated from the global controller  170 , and the control signals RSN, RSP, WSN, WSP, WHSN and WHSP are generated from the local controller  180 . 
     FIG. 14 is a structural diagram illustrating one of a plurality of unit blocks in a cell array block of FIG.  4 . 
     Each unit block comprises a main bitline pull-up controller  330 , a cell array, and a column selection controller  310 . The cell array includes a main bitline load controller  340 , and a plurality of sub cell blocks  350  connected in series between the main bitline pull-up controller  330  and the column selection controller  310 . 
     FIG. 15 is a circuit diagram illustrating the main bitline pull-up controller  330  of FIG.  14 . The main bitline pull-up controller  330  comprises a PMOS transistor having a gate connected to a control signal MBPUC, a source connected to a Vpp or a Vcc, and a drain connected to the main bitline  360 . The main bitline pull-up controller  330  pulls up the main bitline to a “high” level in a “precharge” operation. 
     FIG. 16 is a circuit diagram illustrating the column selection controller  310  of FIG.  14 . The column selection controller  310  comprises a transmission gate for connecting a main bitline to a data bus line in response to control signals CSN and CSP. 
     FIG. 17 is a circuit diagram illustrating the main bitline load controller  340  and the sub cell block  350  of FIG.  14 . Here, one sub cell block  350  is shown for convenience sake. The main bitline load controller  340  comprises a PMOS transistor having a gate connected to a control signal MBLC, a source connected to a Vpp or a Vcc, and a drain connected to the main bitline  360 . 
     When the control signal MBLC is activated, the main bitline load controller  340  serves as load of the main bitline  360 . A detection voltage of the main bitline  360  is determined by a load resistance and a current level of the main bitline  360 . The current level is determined by a transistor N 1 . The main bitline load controller  340  may be attached to each main bitline. However, when a driving load is large, the main bitline load controller  340  is arranged in each sub cell block  350 , thereby reducing driving load of each main bitline load controller  340 . 
     The sub cell block  350  comprises a sub bitline  351 , and NMOS transistors N 1 , N 2 , N 3 , N 4  and N 5 . The sub bitline  351  is connected in common to a plurality of unit memory cells. Each unit memory cell is connected between a wordline WL&lt;m&gt; and a plateline PL&lt;m&gt;. The NMOS transistor N 1  for regulating current has a gate connected to a first terminal of the sub bitline  351 , and a drain connected to the main bitline  360 . The NMOS transistor N 2  has a gate connected to a control signal MBSW, a drain connected to a source of the NMOS transistor N 1 , and a source connected to ground. The NMOS transistor N 3  has a gate connected to a control signal SBPD, a drain connected to a second terminal of the sub bitline  351 , and a source connected to ground. The NMOS transistor N 4  has a gate connected to a control signal SBSW 2 , a source connected to the second terminal of the sub bitline  351 , and a drain connected to a control signal SBPU. The NMOS transistor N 5  has a gate connected to a control signal SBSW 1 , a drain connected to the main bitline  360 , and a source connected to the second terminal of the sub bitline  351 . 
     The load of the main bitline may be reduced to that of the sub bitline  351  by activating one of a plurality of sub bitlines  351  in the main bitline  360 . The sub bitline  351  is selected by the control signal SBSW 1 . 
     The sub bitline  351  regulates a potential of the sub bitline  351  to a ground level if the SBPD signal, which is a regulating signal of the pull-down NMOS transistor N 3 , is activated. 
     The SBPU signal regulates a power voltage to be supplied to the sub bitline  351 . When a “high” voltage is required in a low voltage, a voltage higher than the VCc voltage is supplied to the sub bitline  351 . 
     The control signal SBSW 2  regulates a signal flow between the sub bitline SBL and the main bitline MBL. The sub bitline  351  is connected to a plurality of unit cells. 
     The sub bitline  351  is configured to be connected to the gate of the NMOS transistor N 1  and to regulating a sensing voltage of the main bitline  360 . 
     FIG. 18 a  is a timing diagram illustrating a write operation of the sub cell block of FIG.  17 . 
     In intervals t 2  and t 3 , a level of a signal written in a cell is detected. In an interval t 4 , a self-boosting operation is prepared. In an interval t 5 , a “high” level signal is written. In an interval t 6 , a “low” level signal is written. 
     In the intervals t 2  and t 3 , if data of the cell is “high”, a voltage of the sub bitline  351  becomes “high”. As a result, as current flowing in the NMOS transistor N 1  becomes larger, a voltage of the main bitline  360  becomes lower than the reference level. On the other hand, when the data of the cell is “low”, the voltage of the sub bitline  351  becomes “low”. As a result, as the current flowing in the NMOS transistor N 1  becomes less, the voltage of the main bitline  360  becomes higher than the reference level. In this way, the data stored in the cell may be detected. 
     In the interval t 4 , if the SBSW 2  becomes “high” at a state where the SBPU is maintained at a “low” level, charges are stored in parasitic capacitors between the gate and the source or the gate and the drain of the transistor N 4 . In the interval t 5 , if the SBPU becomes “high”, potentials of the SBSW 2 , the sub bitline  351  and the wordline WL&lt;I&gt; are boosted as much as additional potential difference by the stored charges. As a result, data “1” is automatically stored in the cell. 
     If the data inputted to the main bitline  360  through the I/O buffer is “0”, the SBSW 1  is activated, and the SBSW 2  is inactivated. Then, the potential of the plateline PL&lt;i&gt; becomes “high”, and that of the sub bitline  351  also becomes “0”. As a result, as the charges stored in the cell move into the sub bitline, the data “0” is written in the cell (t 6 ). 
     FIG. 18 b  is a timing diagram illustrating a read operation of the sub cell block of FIG.  17 . 
     In intervals t 2  and t 3 , a level of a signal written in a cell is detected. In an interval t 5 , a “high” level signal is written. In an interval t 6 , a “0” level signal is restored. 
     The operation in the intervals t 2 -t 4  is identical with that of FIG. 18 a . In general, a restore operation is required after a read operation. Referring to FIG. 18 b , however, a restore operation is performed in the intervals t 5  and t 6 . In the interval t 5 , data “1” is restored regardless of the originally stored value. In the interval t 6 , data “0” is restored. The explanation of the restore operation is omitted because it is the same as that of the write operation. 
     FIG. 19 is a cross-sectional diagram illustrating the connection portion between a data bus unit and a column selection controller according to the present invention. The connection portion comprises a first layer L 1 , a second layer L 2 , and a third layer L 3 . The first layer L 1  comprises two NMOS transistors having a common source and a common drain. The common source is connected to the main bitline  360 , and the common drain is connected to a first shared layer  370 . The second layer L 2  includes a second shared layer for connecting the first shared layer  370  to the data bus line  210 . The third layer L 3  includes the data bus line  210 . 
     The second shared layer  380  allows the area of the first shared layer  370  to be minimized. As a result, the increase in the area of the whole chip layout due to increase in that of the first shared layer  370  can be prevented. Additionally, the design of the above-described layout improves process margin and efficiency of signal transmission. 
     FIG. 20 is a block diagram illustrating a VPP supply circuit  700  for supplying a VPP to the cell array block of FIG.  4 . Referring to FIG. 20, a plurality of VPP driving circuits  800  are arranged in each cell array block  300 , and first VPP pump circuits  200  are arranged on top and bottom of the control circuit unit  100 . 
     The VPP driving circuit  800  comprises a second VPP pump circuit  820 , a level shifter  810 , and a driver  830 . The second VPP pump circuit  820  generates a gate VPP signal. The level shifter  810  level-shifts an output signal from the first VPP pump circuit. The driver  830 , which is controlled by the gate VPP signal and an output signal from the level shifter  819 , outputs a driving voltage. 
     The first VPP pump circuit  700  requires relatively larger layout size and operates at lower speed. Therefore, the first VPP pump circuit  700  is disposed in the middle of the VPP driving circuits  800  to effectively control a VPP level. However, the VPP driving circuit  800  operates at higher speed. As a result, the VPP driving circuit  800  is individually arranged in each unit block. 
     FIG. 21 is a structural diagram illustrating the VPP driving circuit  800  of FIG.  20 . 
     The driver  830  comprises NMOS transistors  821  and  832 . The NMOS transistor  832  has a source connected to ground, a drain to output a driving voltage, and a gate controlled by a result obtained from logical operation of a pull-down control signal and an output signal from the address decoder. The NMOS transistor  821  has a gate to receive the VPP (gate Vpp signal) supplied from the second VPP pump circuit  820 , a source to receive the output signal from the address decoder, and a gate connected to a drain of a NMOS transistor  831  in a node N 1 . 
     A level shifter circuit comprises the NMOS transistor  831 . The NMOS transistor  831  has a drain to receive the VPP (driving Vpp signal) outputted from the first VPP pump circuit, a source connected to a drain of the NMOS transistor  832 , and the gate connected to the drain of the NMOS transistor  821 . 
     FIG. 22 is a timing diagram illustrating the operation of the VPP driving circuit  800  of FIG.  21 . In the cell operation, the voltage VPP is outputted into the wordline, the plateline, the SBPU and SBSW 2  (see FIG.  17 ). Referring to FIG. 22, in an interval T 1 , a signal WLCON as an output signal from the second VPP pump circuit  820  is to make the node N 1  (see FIG. 21) at a VCC level. If a gate voltage of the NMOS transistor  821  rises to the VPP, a voltage smaller than the VPP is generated at the node N 1 . This value is represented by a VCC. Then, if the gate voltage of the NMOS transistor  821  becomes VCC, the NMOS transistor  821  is turned off and the node N 1  becomes floating. 
     When an interval T 2  starts, a driving VPP signal outputted from the first VPP pump circuit  700  becomes VPP. Here, the node N 1  is boosted to VCC+VPP by charges stored in the parasitic capacitor between the node N 1  and the drain of the NMOS transistor  831 . Additionally, a wordline voltage becomes VPP because the NMOS transistor  831  is turned on. 
     The VPP of the WLCON signal in the final stage of the interval T 2  turns on the NMOS transistor  821 . As a result, the node N 1  falls to the VCC level. Here, since the WLCON signal falls to the VCC again, the NMOS transistor  821  is turned off and the node N 1  becomes floating. The voltage of the node N 1  also becomes the VCC, and the NMOS transistor  831  is turned off. As a result, the wordline WL becomes floating. Here, a potential difference is maintained by charges stored in the parasitic capacitor between the floating wordline WL and the sub bitline  351 . In this state, if the SBPU signal becomes “high”, the sub bitline  351  becomes “high” (see FIGS.  17  and  18 ). Next, a boosting voltage is generated as much as the potential difference maintained between the wordline WL and the sub bitline  351  (T 3 ).