Abstract:
A time division multiplex data recovery system using a closed-loop phase lock loop (PLL) and delay locked loop (DLL) is disclosed. In other words, one closed loop comprises both a phase locked loop (PLL) and a delay locked loop (DLL) in a novel time division multiplex data recovery system. This new architecture comprises a 4 stage Voltage Controlled Oscillator (VCO) used to generate 8 clock signals, 45 degrees phase shifted from one another, for 8 receivers to do the oversampling. An interpolator tracks the received data signal and feeds it back to the Phase/Frequency Detector (PFD). The PFD has a second input of the reference clock which the PFD uses along with the interpolator input to correct the frequency of the PLL. The PLL operates at a high bandwidth. The DLL&#39;s bandwidth is several orders lower than the PLL. The DLL activates only a multiplexer and an interpolator continuously, thereby drawing a minimum amount of power.

Description:
FIELD OF THE INVENTION 
   This invention relates in general to time division multiplex data recovery systems and more specifically to a time division multiplex data recovery system using a close loop phase and delay locked loop. 
   BACKGROUND OF THE INVENTION 
   The TI Wizard is a full Duplex SERDES (Serialize-Deserialize) function plus the high speed differential I/O, the built-in self test (BIST) circuitry, and the phase locked loop (PLL) and delay locked loop (DLL) circuitry needed for timing references for the SERDES. A Wizard port supports data rates from 1.0 Gbaud through 2.5 Gbaud. It also includes comma detect circuitry used for the byte alignment of incoming bit stream. 
   The Wizard performs basically a Fibre Channel operation: Transmit is 10-bits of 8b/1 0b encoded data in and 1 bit at 10× the frequency differential out Receive is a 10× speed serial bit stream in and 10-bit 8b/1 0b data out. The receive side also performs byte alignment. Wizard is also intended to be used as the Physical Interface Portion of the 800 Mb 1394 standard, as well as Gigabit Ethernet, board to Board interconnect, DSP test port interface, chip to chip interconnect, and any application that requires a very high speed serial interface. The primary intended application for Wizard is to provide building blocks in the ASIC library for developing point-to-point baseband data transmission over controlled impedance media of approximately 5052. The transmission media can be printed circuit boards, copper cables or fiber-optical media. The ultimate rate and distance of data transfer is dependent upon the attenuation characteristics of the media and the noise coupling to the environment. 
   A Wizard port performs the data serialization, deserialization, and clock extraction functions for a physical layer interface device. The transceiver operates at a maximum of 2.5 Gbps of bandwidth over a serial interface. The transmitter portion serializes 10-bit data words at a rate based on the supplied reference clock. The 10-bit word is intended for 8b/1 0b encoded data, which is then serialized and transmitted differentially at the intended data rate using Non-Return-to-Zero (NRZ). The receive section deserializes the input data and synchronizes the 10-bit wide bus data to the local reference clock. 
   As shown in  FIG. 1 , a phase locked loop (PLL) is used for clock multiplication and a delay locked loop (DLL) is used for clock recovery. The speed of the reference clock (REFCLK) of the PLL is 100-250 MHz depending upon the ASIC or board speed of the data link. Transmit path registers the parallel data TD&lt; 0 : 9 &gt; at the rising edge of REFCLK, does the parallel to serial conversion and sends the serial data through the transmitter at both edges of the TX clock. Receive path receives serial data and does serial to parallel conversion using the RX clock. 
   One prior art structure of a time division multiplex data recovery system uses one PLL as the reference clock and two open-loop DLLs for data recovery. Although this system has the advantage of being stable in a noisy environment due to the high bandwidth of the PILL and passive DLL clock recovery, this system also has the great disadvantage of requiring high power due to use of the high speed clock which necessitates a huge clock buffer and additional digital logic processing the output of the receiver. Also disadvantageous is that the majority of the system operates digitally thus increasing the power required as the frequency of operation increases. 
   A second prior art system uses a single PLL for data recovery. Although this system has the advantages of operating at high speeds due to the time division multiplex of 20 slow clocks(assuming a 1:10 deserializer), low power operation due to no clock buffer requirement (clock fanout is one, i.e. capacitive load is very small) and few digital circuits in operation, and activity of all the transmitter and receivers is at a 10% duty cycle due to the time division multiplex, it also has the disadvantage of the data loop receiver having large hunting jitter and the system is not stable in noisy environments due to the extremely narrow bandwidth of operation. 
   SUMMARY OF THE INVENTION 
   A time division multiplex data recovery system using a closed-loop phase locked loop (PLL) and a delay locked loop (DLL). In other words, one closed loop comprises both a phase locked loop (PLL) and a delay locked loop (DLL) in a novel time division multiplex data recovery system. This new architecture comprises a 4 stage Voltage Controlled Oscillator (VCO) used to generate 8 clock signals, 45 degrees phase shifted from one another, for 8 receivers to do the oversampling in the data recovery block. The receivers  44  each trigger on only the rising edge of the clock to alleviate duty cycle issues. A phase interpolator  44  tracks the received data signal and feeds it back to the Phase/Frequency Detector (PFD)  34  to adjust the measurement position of the sampling clock  70 . The PFD  34  has a second input of the reference clock which the PFD uses along with the interpolator input to correct the frequency and phase of the PLL. The PLL operates at a high bandwidth. The DLL&#39;s bandwidth is several orders lower than the PLL. The DLL activates only a multiplexer and an interpolator continuously, thereby drawing a minimum amount of power. 

   
     BRIEF DESCRIPTION OF THE DRAWING 
       FIG. 1  illustrates a block diagram of the TI Wizard full Duplex SERDES (Serialize-Deserialize) function. 
       FIG. 2  illustrates a block diagram of a time division multiplex data recovery system of the prior art. 
       FIG. 3  illustrates a block diagram of a time division multiplex data recovery system using a closed-loop phase lock loop(PLL)  30  and delay locked loop(DLL)  32  according to a preferred embodiment of the invention. 
       FIG. 4  illustrates a block diagram of the VCO  38  of the PLL  30  according to a preferred embodiment of the invention. 
       FIG. 5  illustrates an eye diagram of the data recovered by the receivers  41  illustrated in the VCO  38  of FIG.  4 (although in actuality, the receivers  41  are located in the data recovery block  40  of FIG.  3 ). 
       FIG. 6  illustrates the relationship between the sampling clock, the leading clock, the input data, and recovered byte clock and leading byte clock waveforms of the Early/Late Voting logic of the data recovery block of FIG.  3 . 
       FIG. 7  illustrates a waveform diagram of the two data bits having an early/late comparison performed in the Early/Late voting logic of the data recovery block of FIG.  3 . 
       FIG. 8  illustrates the first stage of the early/late bit comparison pipeline of the Early/Late Voting logic of the data recovery block of FIG.  3 . 
       FIG. 9  illustrates the first stage of summing up the early and late votes of the Early/Late voting logic of the data recovery block of FIG.  3 . 
       FIG. 10  illustrates the Early/Late accumulation stage of the Early/Late voting logic of the data recovery block of FIG.  3 . 
       FIG. 11  illustrates the final stage of the Early/Late pipeline of the Early/Late voting logic of the data recovery block of FIG.  3 . 
       FIG. 12  illustrates a simplified phase select block diagram which illustrates the operation of the Phase interpolator  44  of FIG.  3 . 
       FIG. 13  illustrates the 16-bit thermometer code  82  distributed equally between two input clock signals of the Phase interpolator  44  of FIG.  3 . 
       FIG. 14  illustrates the 8 octants of the phase diagram of the 8 states(clock signals) which are coded into the 3-bit octant code of the Phase interpolator  42  of FIG.  3 . 
       FIG. 15  illustrates, in more detail, the block of circuitry in  FIG. 12  used to correct for invalid codes in the Phase interpolator  42  of FIG.  3 . 
       FIG. 16  illustrates a block diagram of the phase interpolator  44  and periphery circuits of FIG.  3 . 
       FIG. 17  illustrates a block diagram ofjust the phase interpolator  44  of FIG.  3  and FIG.  16 . 
       FIG. 18  illustrates the eight 45 degree phase shifted phases from the VCO  38  used for phase interpolation in the phase interpolator  44  of FIG.  3 . 
       FIG. 19  illustrates a schematic of the phase interpolator  44  of  FIGS. 16 ,  17  and  3 . 
   

   DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
   Architecturally, the transceiver uses a conventional PLL  30  frequency synthesizer on the Transmit side to generate high speed timing references. But on the receive side, according to a preferred embodiment of the invention, it uses an interpolator  44  and a bimodal data/clock receiver  40 , 41 , 42  to determine the ideal placement of the theoretical recovered clock. The phase select circuit  42  and phase interpolator  44  then picks one of the many, i.e. 8, phases of the Delay Locked Loop (DLL)  32  (driven by the Transmit PLL  30 ) that best matches the ideal and uses it as the recovered clock. This process continuously repeats itself as the incoming frequency is slightly off-frequency from the reference frequency. 
   The Phase Locked Loop (PLL)  30  design consists of a phase/frequency detector  34 , a differential charge pump, an on chip loop filter capacitor  36 , a replica-feedback current source bias, a differential voltage controlled oscillator (VCO)  38 , differential to single end buffer stages and a divide by five circuit block  46 . The PLL  30  operates at 1.25 GHz, worst case, and minimizes the clock jitter caused by 250 MHz with 250 mV of low frequency square wave supply noise. 
   There are two important parameters in a Phase Locked Loop (PLL)  30  design. They are the damping factor and the loop bandwidth. The damping factor can be simplified as: 
       ζ   =       y   4     ⁢         x   i     N       ⁢         C   1   2         C   B     (       C   1     +     C   2                   
 
   where: y is the ratio of the small signal resistance of loop filter resistance of symmetric load in the delay element in the Voltage Controlled Oscillator (VCO), x is the ratio of charge pump current to the bias current set in the delay element of the VCO, N is the VCO frequency multiplier, C 1  is loop filter capacitance, C B  is the total effective capacitance of VCO ring oscillator and C 2  is the third order-roll off capacitance. The loop bandwidth to operating frequency ratio is given by 
           ω   m       ω   REF       =       xN     2   ⁢   π       ⁢         C   B       C   1               
 
   In the transceiver PILL  30  design, y=3, x=0.5, N=5, C 1 =12 pf, C b =0.372 pf, C 2 =0.6 pf. 
   As seen from left to right in  FIG. 3 , the Phase Frequency Comparator (PFD)  34  will assert equal and short duration pulses at the outputs when both inputs are in phase. These pulses will help eliminate a dead-band in phase to frequency conversion which will lead to additional input tracking jitter. Since the outputs of PFD  34  (UP and DN) are single ended signals, these two signals are converted into differential UPs and DNs signals by the Phase Splitter Buffer or the INV  32  blocks, which are included in the PFD  34  of FIG.  3 . INV  32  blocks generate two differential UPs and two differential DNs to drive two differential charge pumps. The differential UPs and DNs signal from the output of the INV  32  blocks should be matched as close as possible to help achieve zero static phase offset in the differential charge pump. The differential charge pump circuit, included in the loop filter block of  FIG. 3 , is composed of three identical differential pair stages. The two differential input stages reduce noise coupling from full swing UPs and DNs signals to nearly DC signal at the charge pump output. The output differential pair sums the “phase-to-current conversion” currents at the loop filter node. Each differential pair consists of symmetric loads, two NMOS source coupled pair and NMOS current source which is driven by the current replica circuit output Vcn. Since the PLL  30  is capable of operating over a wide range of frequency and the differential charge pump is scaled inversely proportional with the operating frequency, the differential charge pump current is set high (i.e. ˜250 μa) at 2.6 Gbps operating frequency so that it does not go below 10 μa when the operating frequency is low. 
   Loop filter  36  components consist of resistors and capacitors (third order loop) forming a low pass filter and are all on chip components. Capacitors are N-well PMOS devices with gate connected to control voltage node and drain, source and backgate connect to power supply. In a noisy substrate environment, Nwell device is preferred for a noise sensitive circuit. The loop filter capacitor is 12 pf and the third order roll-off capacitor is set approximately to 0.6 pf(equal to 12 pf/20). The third order capacitor is used to help reduce high frequency ripple on the control voltage Vcp. The total capacitance of the third order capacitor should be less than the loop filter capacitor divided by 20 which is limited by the phase margin constraint. 
   The replica feedback bias circuit, located in the loop filter  36  of  FIG. 3 , is used to generate Vcp and Vcn from the loop filter voltage Vctrl for application to the VCO  38  and the phase interpolator  44 . There are two replica-feedback bias circuit blocks. One is used in the core PLL  30  (to bias the VCO  38 ) and the other is used for biasing the phase interpolators  44  of the transmitters and receivers. This scheme helps to keep the closed loop bandwidth of both replica feedback bias circuits high so that they can track power and substrate noises that can effect the PLL  30  jitter performance. Each replica feedback bias circuit consists of an amplifier bias, a common source PMOS differential amplifier, a half buffer replica circuit and a loop filter control Vctrl voltage buffer. The transistors in loop filter  36  control Vctrl voltage buffers are adjusted so that 1/gm of the PMOS transistors is the desired feed-forward zero resistor value. Since the replica-feedback circuits are self-biasing circuits, bias start up circuit is essential to jump start the replica-feedback bias circuits which in turn initialize the PLL circuit properly. 
   The core of the VCO  38  consists of 4 differential delay elements  50 ,  52 ,  54  and  56  connected in a ring (FIG.  4 ). The differential output at each node in the ring is buffered by three more stages of delay element at 1×, 2×, and 3× of the core delay element buffer strength. Each differential delay element consists of an NMOS source coupled pair, a current source NMOS bias by Vcn and a pair symmetric load. Each symmetric load consists of a diode-connected PMOS device in shunt with an equally sized Vcp biased PMOS device. 
   The 8 differential signal outputs from the VCO are converted into 8 phase distinct inputs, separated by 45 degrees, to directly drive the eight receivers  41  of the data recovery block  40 . This is unlike the prior systems which use high speed clock buffers driving the receivers  41 . The 8 phase distinct inputs also form inputs to the phase interpolator  44 . The bandwidth of this buffer should be high so that the fastest start up oscillation frequency of the VCO in the strong process corner still can pass through this buffer to close the feedback loop. A divide by 5 circuit  46  is used to divide the frequency of the signal at the output of the phase interpolator buffer  44 . The final signal is then buffered and looped back to the PFD  34  in a closed loop to provide clock phase information feedback to the VCO  38  In this way, if the feedback clock is late, the charge pump in the PLL  30  will charge the PLL  30  clock faster to move the feedback clock to align with the reference clock. As the charge pump charges the PLL  30  and shifts the phase of the clock of the VCO  38 , all 8 outputs of the VCO  38  will shift in parallel and will shift likewise, in the same amount and in the same direction. 
   The differential sampling receiver  41  design, illustrated in block diagram form in the VCO  38  of  FIG. 4  operates at 1.25 Gbs and is driven by the 8 phase shifted clock signals generated by the VCO  38 . The data recovery portion DLL of the loop of the time division multiplex data recovery system according to a preferred embodiment of the invention, starts with a sampling receiver  41 . This system uses a single edge clocking scheme so there are four sampling receivers rx 0   60 , rx 1   62 , rx 2   64  and rx 3   66 , called data receivers to recover the data from the input data stream—two for the odd bits and two for the even bits. There are also four more sampling receivers, rxd 0   61 , rxd 1   63 , rxd 2   65  and rxd 3   67 , used to extract the phase information to be used, along with the extracted data information, to tune the sampling data triggering edge such that it is triggering in the middle of the data eye when the receive portion of the loop is in lock. Each one of the 8 receivers  41  are driven by one of the 8 phase shifted by 45 degrees signals from the VCO  38 . The data output from the receivers  41 , through the aid of additional circuitry, attains CMOS levels. 
   The early/late voting logic, illustrated in FIG.  8  through  FIG. 11  or more comprehensively FIG.  5  through  FIG. 11 , is present in the data recovery block  40  of FIG.  3 . The purpose of the early/late voting logic is to determine whether the high speed sample clock of the receiver  41  is early or late relative to the ‘ideal’ temporal position of the sampling clock. The high speed clock that is used to sample the deserialized data stream will be referred to as the “sampling clock”  70 . The high speed clock that leads the sampling clock by 90 degrees will be referred to as the ‘leading clock’  72 . The ideal placement of the sampling point would be such that the high speed data is sampled right in the middle of the data bit&#39;s eye opening as illustrated in  FIG. 5   a  where the sampling clock  70  is located. The early/late voting logic uses a high speed clock that leads the high speed sampling clock by 90 degrees, so that the high speed data sampled by this leading clock  72  would be sampled deliberately right in the middle of the data crossings in the eye diagram as illustrated in FIG.  5 . This data, sampled by the leading clock  72 , is compared to the data sampled by the sampling clock  70  to determine whether the sampling clock is early or late. The output of the early/late voting logic controls the phase select logic circuitry  42 . The phase interpolator  44 , early/late voting logic and phase select logic circuitry  42  all comprise a delay locked loop  32 . 
   The deserializer logic is clocked off of the sampling logic. The early/late voting logic is clocked off the voting clock. The deserializer divides down the sampling clock by 5 to derive the recovered byte clock. This byte clock is synchronous to the deserialized data byte, and the recovered byte clock is used to clock whatever logic uses the deserialized data byte, such as the comma detect logic or the PRBSVER logic. The early/late voting logic divides down the leading clock by 4 to derive a leading byte clock, and this byte clock is synchronous to the voting result of the early/late voting block. This leading byte clock is used to clock the phase select logic  42  that uses the voting result of the early/late voting block. This leading byte clock is used to clock the phase select logic  42  that uses the voting result of the early/late voting block.  FIG. 6  illustrates the relationships between these clocks. 
   The early/late voting logic is constructed in a pipelined fashion, two bits at a time. The input clock is the half-speed half baud rate clock, with rising edge leading the sampling clock  70  by 90 degrees. The first function is to determine from the data bits I whether the sampling clock  70  is early or late in the bit period as illustrated in  FIGS. 5 and 7 . Then these early or late indications are summed in pipelined fashion. This sum is accumulated over an 8-bit window. The early sum over an 8 bit window is compared to the sum of the late indications. If there are more early indications, then the early/late voting logic indicates an early vote. If there are more late indications, then the early/late voting logic indicates a late vote. The logic also derives a byte clock from the high speed clock by doing a divide by 4 (as earlier stated). The early/late voting result is output for every 8-bits of the data stream, and is provided synchronous to the early/late output byte clock. 
   The early/late voting logic first compares the two data bits to the corresponding data bits that were sampled with the sampling clock. This comparison determines whether the bit was sampled early or late.  FIG. 7  illustrates the concept. Suppose for the moment that the data pattern is alternating 1&#39;s and 0&#39;s, with no jitter of the edges. If the sampling clock  70  is perfectly located in the middle of the bit, and the leading by 90 degree clock  72 (hereafter called the voting clock  72 ) is truly leading by 90 degrees, then the voting clock  72  will deliberately be sampling the data bits during their transition. If the sampling clock  70  were to drift early in the bit time, then the voting clock  72  would also tend to drift early. If the voting clock  72  were early, then the voting clock  72  would actually be sampling the previous bit. In this case, an “XOR” of the two bits would reveal that the two bits were different, which would be taken as an indication of the clocks being early. If the sampling clock were to drift late into the bit time, then the voting logic would also tend to drift late. If the voting logic were late, then the voting clock would actually be sampling the same bit as the sampling clock. In this case, an XOR of the two bits would indicate that the bits were the same, which would also be taken as an indication of the clocks being late. 
     FIG. 8  illustrates the first stage of the early/late pipeline, where the data bit sampled with the sampling clock  70  is compared to the data bit sampled with the leading clock  72 . PHINO and PHIN 1  are the data sampled by the leading  72  or voting clock  72 , while RXBITINO and RXBITIN 1  are the data sampled by the sampling clock  70 . An XOR gate is all that is needed to do the compare. Because of the half-baud rate clocks, the pipeline compares two bits at a time. 
   It should be noted that this early/late comparison is only valid if the stream has a logic transition from the current bit used for voting and the previous bit. If the data stream does not transition, then the comparison to the previous bit will indicate a ‘late’ result no matter whether the clocks were late or early because the bits will always be the same in the comparison. If the data stream had two ‘1s’ in a row or two ‘0s’ in a row, then a comparison of the two bits would always yield a ‘same’ result, even if the clocks were not ‘late’ in the baud interval. Thus, when there is not a data transition on the data stream, the voting logic must be inhibited. An XOR gate is needed to compare a data bit with the preceding data bit, and this signal is used to ‘mask out’ the early/late voting comparison. The mask comparison is done in the deserializer logic, and the mask signal is passed as an input to the early/late voting block.  FIG. 9  illustrates the masking stage of the early/late pipeline. There will effectively be an additional vote besides an ‘early’ vote and a ‘late’ vote—there will also be a ‘neither’ vote. Thus, in every 8-bit window of summing votes, the sum of early votes and late votes will not total 8 unless there are data transitions within that 8-bit window of data. 
     FIG. 10  also illustrates the first stage of summing up the early and late votes. In addition to gating the ‘difference’ comparison from  FIG. 9  with the mask bit, the stage in  FIG. 10  sums the votes (since we are working on two bits at a time) with what are essentially half adders. The output ‘Either early’ indicates that either one or the other of the two votes are ‘early’. The output of ‘Both early’ indicates that both of the two votes are ‘early’. The output of ‘Either late’ indicates that both of the two votes are ‘late’. Because there may be less than two data transitions present, the sum of the ‘either’ and ‘both’ votes may not add up to two votes present. In fact, if there are no data transitions, all outputs from  FIG. 9  will be inactive. 
   After the circuit of  FIG. 9 , the sum of early and late votes over a two-bit window of the data stream is available. The next stage is used to sum the votes over an 8-bit window of the data stream. This is done in the circuit loosely illustrated in FIG.  10 . The outputs of  FIG. 10  are summations of the outputs of FIG.  8 . The output ‘Late 1 ’ indicates that there was AT LEAST one late vote over the 8-bit window. The output ‘Late 2 ’ indicates that there were at least two late votes over the 8-bit window. The rest of the outputs are named accordingly. It is only necessary to calculate vote accumulations for up to 3 or more ‘Late’ votes and up to 4 or more ‘Early’ votes. These 7 signals are enough to determine whether there are more late votes than early votes, or vice versa. 
   Each of the flip flops of  FIG. 10  are essentially an accumulate and hold circuit. Any ‘Either Late’ vote will set the ‘Late 1 ’ flip flop, and this value will be held. Also, any ‘Both Late’ indication will set the ‘Late 2 ’ flip flop, directly. Any ‘Either Late’ vote when there already is a ‘Late 1 ’ indication will set the ‘Late 2 ’ flip flop, and this value will be held. Any ‘Either Late’ vote, when there already is a ‘Late 2 ’ indication will set the ‘Later flip flop. Thus, after 4 clock cycles the accumulator in  FIG. 10  is reset to zero to start a new summation. This form of accumulation was chosen because it could be implemented in two stages of three-input logic, and is probably (along with the circuit of  FIG. 11 ) the speed limitation of all the Wizard digital logic. 
     FIG. 11  illustrates the final stage of the Early/Late pipeline. Given the outputs of  FIG. 10 , one can now determine whether there are more early votes, more late than early or the same number of early and late votes. If there is at least one late vote, and not two early votes, then the result is a majority of late votes. A one-to-one tie is taken as a late majority. Likewise, at least two late votes and not three early votes lead to a late majority and at least three late votes and not four early votes lead to a late majority. A two-to-two tie and a three-to-three tie are also taken as a late majority. All other cases are taken as an early majority. Thus, a zero-to-zero tie, and a four-to four tie are taken as an early majority. 
   The reason that a tie is not allowed is that the phase select logic that adjusts the phase of the high speed clocks must always adjust the phase one step earlier(if there is a late majority), or one step later (if there is an early majority). There is no provision for staying on the same phase step. Thus, a tie vote must be biased one way or the other. It was chosen that some tie votes would be taken as early, and the others taken as late to both minimize logic and to prevent the sample clock  70  from being systematically biased towards one side or the other of the ideal sampling point. 
   The comparison circuit of  FIG. 11  uses the clear pulse that is generated every four clock cycles to multiplex between generating a new vote or holding the previous vote. This holds the early/late vote stable for a whole 8 bit time period so that the logic to follow has adequate set-up and hold provision between the EARLYB signal and the CLKOUT signal. The early/late voting output is defined such that a low signal indicates an early vote and a high signal indicates a not early, or late vote. 
   The logic of FIGS.  10 . and  11  are among the most complex of the high speed digital logic. For this reason they were both implemented with a variant of pre-charged dynamic logic that maximizes the use of the faster n-channel transistors for the critical paths of the logic. This variant of dynamic logic is not dependent on the clock rate being above a certain rate, so there is no danger of charge bleeding off at slow or even near DC speeds. 
   Noteworthy is that the leading phase clock  72  is deliberately sampling the incoming data right at the ideal transition point. This is not normally the desired operation. The incoming data is being deliberately sampled right at the ideal data transition point. This is deliberately violating stable set-up and hold conditions on the flip flops that the data is being clocked into, leading to potentially unstable conditions. In reality, the data transitions will jitter around the ideal edge placement in practice so many times the data will be sampled accurately with the leading phase clock. On many other occasions this will not be so, however. The rationale is that on those occasions that the leading edge does catch the data right on transition, it does not matter whether the data was caught correctly or not. In this case, the metastable sampling of the data indicates that the leading edge clock is precisely positioned where it should be anyway. Since the phase select logic  42  MUST take a phase step early or late, either one will be a step away from ideal, and it does not matter which is chosen. It is expected that on the next phase update, the phase select logic  42  will step back towards the ideal sampling point again. Thus, there is a measure of built-in jitter on the recovered clock. The only remaining concern is that the early/late voting logic and the phase select logic  42  be implemented in such a way as to not get into an invalid or damaging state in the presence of metastable logic conditions internally. Due to the pipelined nature of the early/late voting logic, metastable logic conditions will not throw the logic into an invalid state and the depth of the pipelined logic actually helps to provide settling time for any metastable logic condition to settle down before it can propagate through to the phase select logic. The remaining concern is that the Phase Select Logic Block  42  can be constructed such that it is tolerant of potentially metastable conditions on the early/late voting result. That topic will be covered in the documentation for the phase select logic  42 . 
   Looking at the bigger picture of the digital locked loop (DLL)  32 , the time required to determine a vote, update the phase select logic  42 , and let the phase interpolators  44  settle to a stable phase all determine the maximum update rate of the DLL  32 . The maximum update rate of the DLL  32  effectively determines the DLL&#39;s  32  bandwidth and its ability to track frequency offset and wander. In the current implementation of the transceiver, the early/late vote takes about 18-bit times to calculate. Allowing 4-bit times for the phase select  42  update and 10-bit times for the phase interpolators  44  to settle, makes for a maximum phase update every 32 bit times. Since there are 64 phase steps per bit time, a maximum tracking rate of {fraction (1/32)} times {fraction (1/64)}={fraction (1/2048)} or 488 parts per million is achieved. 
   The function of the Phase Select block  42  is to generate the digital control word that sets the phase of the recovered clock via the phase interpolator  44 . The phase control word is a 19-bit digital word that selects one of 128 possible phases through the phase interpolators from the analog PLL  30 . The 19 bit control word that selects one of 128 possible phases through the phase interpolators from the analog PLL. The 19-bit I control word comes from a bank of 19 flip flops that are operated as a form of state machine. The Phase Select state machine  42  is clocked from the recovered byte clock from the Early/Late Voting block, which is the high speed leading phase clock  72  divided by 4. On every clock rising edge, the phase control word is updated to point to the next earlier phase, next later phase, or remain on the same phase. 
     FIG. 12  illustrates the operation of the state machine of Phase Select block  42 . A bank of 16 flip flops contain a 16-bit ‘thermometer code’  82  that controls the interpolation section of the phase interpolators  44 . An additional three flip flops hold a 3 bit code that selects the multiplexing of the 8 clock phases into the phase interpolators  44 . The total 19-bit register is operated as a state machine, where each state in the state machine determines the next state, depending upon the value of the EARLYB voting result from the Early/Late Voting block. 
     FIG. 12  shows a divide-by-4 circuit that divides down the input clock to derive a control pulse that goes to multiplexers in front of the flip flops. This is called the decimation circuit and makes certain that the phase state is NOT updated on three clock cycles out of every four. Every fourth clock cycle the phase state is updated to point to the next later state (called the ‘up’ state) or the next earlier state (called the ‘down’ state). The decimation ensures that the phase is only updated every 32-bits, giving the phase interpolators  44  time to settle out before the next early/late voting is taken. 
   On the clock cycle where the phase is updated, the only choices for the new phase is the ‘up’ phase or the ‘down’ phase, depending upon the state of the EARLYB input. The EARLYB input controls a multiplexer that selects the new state, with a high level selecting the down (earlier) state and a low level selecting the up (later) state. The 2-input multiplexers controlled by the EARLYB signal are placed closest to the state flip flops in order to minimize set-up/hold requirements on the EARLYB signal relative to the clock input. Fewer multiplexers would be needed if the multiplexers controlled by the decimation phase in  FIG. 12  was placed closer to the flip flops, and the EARLYB signal then controlled one set of multiplexers that selected between up and down states, but the extra multipliers were deemed worthy for the purposes of keeping the EARLYB set-up/hold requirements down. 
   On each clock cycle of the state machine, the current phase state is fed back through the phase select logic  42  and both the next ‘up’ state and the next ‘down’ state are computed in parallel. Only one of the possible new states are used, depending on the control of the multiplexer, and the other possible new state is discarded and the current state is kept 
   The 16-bit thermometer code  82  is used to select one of 17 possible phases between two input clocks. If the thermometer code  82  is 0000000000000000 then the resultant phase is 100% of the input clock and 0% of the second input clock. If the thermometer code  82  is 1000000000000000 then the resultant phase is −94% of the first input clock and −6% of the second input clock. If the thermometer code  82  is 1111111111111111 then the resultant phase is 0% of the first input clock and 100% of the second input clock. See  FIG. 13  for illustration. At the code 1111111111111111, 0% of the first input clock is used to determine the resultant phase, and so at this code the mux input to the phase interpolator may by switched over to a third input clock with no change in the resultant phase. If the thermometer code  82  then becomes 1111111111111110, then the resultant phase is 94% of the second input clock and 6% of the third input clock. Thus the name thermometer code  82 : the code rises from 0000000000000000 to 1111111111111111 as the phase interpolator is bumped ‘up’ in phase until the muxes for the input phases are switched, and then the thermometer code  82  falls from 1111111111111111 down to 0000000000000000 as the phase continues to be bumped ‘up’. thus in even numbered octants of the phase diagram (see FIG.  14 ), a rising thermometer code  82  equates to moving ‘up’ in phase, while in odd numbered octants, a failing thermometer code  82  equates to moving ‘down’ in phase. This thermometer code is forwarded to the phase interpolator  44  for adjusting the phase of the sampling clock  70  accordingly. 
   There are 8 octants on the phase diagram of FIG.  14  and these are coded into the 8 states of the 3-bit octant code. The 8 codes were chosen as the sequence of a Gray-code counter so that as the phase moves from one octant to the next in either direction, only one bit of the code changes at a time. This simplifies the coding for the phase interpolator muxs  80 (illustrated in  FIG. 16 ) and also protects against an unstable EARLYB signal from the early/late voting block from being able to ever make the phase jump more than one or two phase steps. This is part of the protection designed to protect against metastability causing disruptive phase steps. 
   The 16-bit thermometer code  82  is also designed so that only one bit at a time should change from one phase step to the next. In this case, instead of the Grey-code pattern, a single next bit in the sequence changes from 0 to 1 or 1 to a 0 such that the code starts off with n ‘1’s and then finishes with 16- n ‘ 0’s. Examples of valid thermometer codes  82  are 1111111100000000 or 1111100000000000. Examples of invalid codes are 0000000000001111 or 1110111000000100. The first is invalid because it has the zeros before the ones while the second is invalid because it has ones after zeros. 
   There is block of circuitry in  FIG. 12  to correct for invalid codes. This circuitry is shown in more detail in FIG.  15 . Essentially, each bit of the code is ANDed with the bit preceding it in the code. If a one in the code is either a first bit of the code or follows a one bit in the code, then it is a valid code. If a one in the code is found to follow a zero, then that one is changed to a zero and then filtered out. In this way, if any invalid code, either from the initial power up states of the flip flops or due to circuit glitch or metastable EARLYB signal, is filtered out back to valid code. 
   The Phase Interpolator  44  illustrated in  FIGS. 3 ,  16 ,  17 , and  19  receives two input clocks with a phase difference of 45 degrees from the VCO  38  and generates an output clock whose phase is the weighted sum of the phases of the two input clocks. It is designed to have 16 phase steps per 45 degrees (64 phase steps per bit time) as illustrated in  FIGS. 13 and 18 . The thermometer code  82  that controls the digital weights is generated by the phase select logic  42 . As shown in  FIG. 4 , the four stage VCO  38  (delay elements are differential) produces 8 clocks which are 45 degrees apart. The phase interpolator  44  takes two adjacent phases and interpolates them to one of the 16 phases based on the digital weights. Two clock phases are selected by the phase select logic  42  and then interpolated by the interpolator based on the thermometer code ic( 0 : 15 ) and ic_b( 0 : 15 )  82 , which are also generated by the phase select logic  42 . 
   As shown in  FIG. 4 , the four stage VCO  38 (delay elements are differential) produces eight clock phases, ck 0 , ck 1 , ck 2 , ck 3 , ck 4 , ck 5 , ck 6 , ck 7 , which are each 45 degrees apart from one another. The interpolator takes two adjacent phases and interpolates them to one of the 16 phases based on the digital weight controlled by the thermometer code  82 . As shown in  FIG. 16 , two MUXes  80  select two adjacent vectors for interpolation. Top PHMUX  80  selects one vector out of four (ck 0 , ck 2 , ck 4  and ck 6 ) and bottom PHMUx  80  selects one vector out of the four remaining vectors Since the signal path is differential, if ck 0  is selected (as the positive signal of the differential ck 4  is the negative signal of the differential signal and vice versa. Both PHMUX  80  are identical and each comprise two more identical multiplexers within each PHMUX  80 . The first multiplexer within selects either ck 0  or ck 2  and the later selects either ck 4  or ck 5  by simply inverting the output of the first MUX. Phase select logic  42  generates control signals sE 0 , sE 1 , sO 0 , sO 1 , invE, invE_b, invO and invO_b, which determine which clock vectors are selected in such a way that only one vector is selected at one instance. For example, if the output clock phase is starting from 0 degrees and increasing (counter clockwise direction in FIG.  18 ), ck 0  and ck 1  are the two vectors selected. When through thermometer inputs demanding an increasing phase shift the output clock reaches 45 degrees, ck 2  is selected instead of ck 0  while ck 1  is still selected. Similarly, when the output clock reaches 90 degrees, ck 3  is selected instead of ck 1  while ck 2  is still selected. The two selected clock phases, oep and oop in  FIG. 16  are interpolated by phase interpolator  44  based on the thermometer code ic( 0 : 15 ) and ic_b( 0 : 15 )  82 , which are generated by the phase select logic  42 .