Abstract:
A reception quality estimation device includes: a correlation extractor for extracting correlation between multiplexed reception reference signals received from the plurality of user terminals and a predetermined reference signal sequence, wherein extracted correlation corresponds to each user terminal; a received power calculator for calculating received signal power from the extracted correlation corresponding to each user terminal; a total received power calculator for calculating total received signal power by summing up the received signal power for all the plurality of user terminals; a noise power estimator for estimating noise power based on the total received signal power and total received power of multiplexed reception reference signals; and a reception quality calculator for calculating the reception quality of each user terminal based on the received signal power and the noise power.

Description:
BACKGROUND OF THE INVENTION 
       [0001]    1. Field of the Invention 
         [0002]    This application is based upon and claims the benefit of priority from Japanese Patent Application No. 2007-230947, filed on Sep. 6, 2007, the disclosure of which is incorporated herein in its entirety by reference. 
         [0003]    The present invention relates to a radio communications system and, more particularly, to a method and device for estimating reception quality. 
         [0004]    2. Description of the Related Art 
         [0005]    The 3rd Generation Partnership Project (3GPP), which is a collaboration between organizations aiming to standardize radio communications systems, has been studying Long Term Evolution (LTE), which provides a high-data-rate, low-latency, and packet-optimized radio-access technology, as a successor to the current W-CDMA systems. In LTE, single-carrier transmission is adopted as the uplink access scheme in broadband radio access. With low peak to average power ratio (PAPR), the single-carrier transmission is excellent in power efficiency, compared with multi-carrier transmission such as Orthogonal Frequency Division Multiplexing (OFDM). Hence, the single-carrier transmission is an access scheme suitable for the uplink from a user terminal (also referred to as UE (user equipment)), which has limited battery capacity, to a base station (also referred to as “Node B” or “eNB”). 
         [0006]    Moreover, for uplink reference signal sequences, Constant Amplitude Zero Auto-Correlation (CAZAC) sequences are used. The CAZAC sequences are sequences having constant amplitude in time domain as well as in the frequency domain and also having zero autocorrelation except when the phase difference is zero. Because of the constant amplitude in time domain, the CAZAC sequences can achieve low PAPR, and because of the constant amplitude in the frequency domain, the CAZAC sequences are suitable for frequency-domain channel estimation. As an example of the CAZAC sequences, Zadoff-Chu sequence (hereinafter, referred to as “ZC sequence”) can be cited, which are represented by the following equation (see 3GPP TS36.211 v1.2.1): 
         [0000]        x   k ( n )=exp(− jπkn[n+ 1 ]/L ) 
         [0007]    In time domain, CAZAC sequences are represented by the following equation (see Popovic, B. M., “Generalized Chirp-Like Polyphase Sequences with Optimum Correlation Properties,” IEEE Transactions on Information Theory, July 1992, Vol. 38, No. 4, pp. 1406-1409): 
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         [0000]    where L is the sequence length of a CAZAC sequence, n is an element number (0, 1, . . . , L-1) in the CAZAC sequence, and k is a sequence number of the CAZAC sequence and is an integer prime to L. The number of CAZAC sequences is dependent on the length of the CAZAC sequences. In the case of the above-mentioned ZC sequence, when the sequence length L is a prime number, the number of sequences is L-1, which is the maximum. That is, the shorter the sequence length, the smaller the number of different CAZAC sequences. 
         [0008]    When CAZAC sequences are used for uplink reference signal sequences, Code Division Multiplexing (CDM) is used to multiplex the reference signals of multiple mobile stations (see 3GPP R1-060925, Texas Instruments, “Comparison of Proposed Uplink Pilot Structures For SC-OFDMA,” March 2006). In CDM of reference signals, users use CAZAC sequences of the same length respectively, and orthogonality between the reference signals can be accomplished by a cyclic shift unique to each user (or antenna). Hereinafter, a brief description will be given of cyclic shifts. 
       1) Cyclic Shift 
       [0009]      FIG. 1  is a schematic diagram to describe cyclic shifts based on a CAZAC sequence. Referring to  FIG. 1 , assuming that a CAZAC sequence C 1  is a sequence  1 , a sequence  2  is generated by shifting the sequence  1  rightward and relocating the shifted-out part at the end of the sequence  1  to the top of the sequence  1 . Moreover, a sequence  3  is generated by shifting the sequence  2  rightward and relocating the shifted-out part at the end of the sequence  2  to the top of the sequence  2 . By sequentially shifting the sequence in a ring manner as described above, sequences  4 ,  5  and  6  are generated. This is called cyclic shifts, and CAZAC sequences generated by cyclic shifts are referred to as cyclic-shift sequences. Hereinafter, cyclic-shift sequences will be represented by S 1 , S 2  and so on by using numbers that indicate shifted amounts. 
         [0010]    Since the autocorrelation value of CAZAC sequences is always zero except when the phase difference is zero as mentioned above, orthogonality between multiple reference signals can be accomplished even in a multi-path environment by making the amount of a cyclic shift to be relocated from the end of a sequence to the top thereof equal to or larger than a supposed maximum delay path time. For example, in a propagation path model according to LTE, since the maximum delay path time is approximately 5 μsec and a single long block is 66.6 μsec long, it is possible to use, logically, 13 cyclic-shift sequences from the calculation of 66.6/5. However, it is thought that approximately six cyclic-shift sequences can be orthogonalized in actuality because an impulse response is broadened along a path due to the influence of a filter and the like (see 3GPP R1-071294, Qualcomm Europe, “Link Analysis and Multiplexing Capability for CQI Transmission,” March 2007). 
       2) Reference Signal 
       [0011]    In LTE, reference signals (hereinafter, abbreviated as “RS” where appropriate) for the uplink can be broadly classified into three types: reference signals for demodulating Physical Uplink Shared Channel (PUSCH), which mainly transmits data, reference signals for demodulating Physical Uplink Control Channel (PUCCH), which transmits a control signal, and reference signals for measuring uplink channel quality or reference signals for CQI measurement (hereinafter, referred to as “sounding reference signal”). 
         [0012]      FIG. 2A  is a format diagram showing an example of resource allocation in a slot including PUSCH and PUCCH, demodulation reference signals for PUSCH and PUCCH, and a sounding reference signal. One slot is composed of seven blocks. Resource blocks (RB) on the edges of the entire band are allocated to PUCCH. PUCCH and PUSCH are multiplexed by Frequency Division Multiplexing (FDM). Additionally, one resource block includes 12 sub-carriers. 
         [0013]    Moreover, PUCCH and the demodulation reference signal for PUCCH, as well as PUSCH and the demodulation reference signal for PUSCH, are multiplexed by Time Division Multiplexing (TDM) in their respective bands. The sounding reference signal is assigned a resource using the system band, independently of the demodulation reference signals for PUCCH and PUSCH. 
         [0014]    In PUCCH transmission as shown in  FIG. 2A , to obtain a larger frequency diversity effect, it is defined in standardization to use CDM to multiplex users spreading over the PUCCH band. In this event, orthogonality between the users can be accomplished as in the above-described CDM of reference signals, by employing CAZAC sequences for the sequences to be used as spreading codes. 
         [0015]    Moreover, in multiplexing of user&#39;s reference signals for PUCCH demodulation, CDM is also used because a number of CAZAC sequences can be secured without a reduction in the sequence length of the reference signals. However, PUCCH needs to be used for not only a user terminal (UE) that has received downlink data to transmit ACK/NACK but also a user terminal (UE) waiting for scheduled downlink data transmission to transmit CQI, which indicates the quality of a downlink for the user. Accordingly, a reference signal for PUCCH demodulation is separated into ACK/NACK use and CQI use. 
       3) Generation of Reference Signal Sequence 
       [0016]      FIG. 2B  is a block diagram showing a functional configuration of a reference signal sequence generation circuit. As described above, a PUCCH control signal is spread by using a CAZAC sequence in frequency domain. First, a sub-carrier mapping section  11  maps a frequency-domain CAZAC sequence to a predetermined sub-carrier, which is then transformed into a time-domain signal by an inverse fast Fourier transform (IFFT) section  12 . Subsequently, a cyclic shift section  13  applies to the time-domain CAZAC sequence signal a cyclic shift of one of the six patterns shown in  FIG. 1 , for example, that is assigned to the user terminal in question. A cyclic prefix insertion section  14  adds a cyclic prefix (CP) to the thus-obtained time-domain CAZAC sequence signal, whereby a reference signal is generated. 
       4) Estimation of Reception Quality 
       [0017]    For channel estimation for multiple user terminals UEs multiplexed by CDM, a frequency-domain cross-correlation method can be used (see,  FIG. 2  in 3GPP R1-070359 NEC Group, “Definition of Cyclic Shift in Code Division Multiplexing,” January 2007). As an example, a description will be given of channel estimation for four user terminals UE 1  to UE 4 . 
         [0018]      FIG. 3A  is a block diagram showing a basic configuration of a multi-user channel estimation device. Referring to  FIG. 3A , after a CP deletion section  20  deletes cyclic prefixes (CP) from received signals to output reference signals, a fast Fourier transform (FFT) section  21  transforms the reference signals into frequency-domain signals. Subsequently, a multiplication processing section  22  carries out complex multiplication of the frequency-domain signals with a single CAZAC sequence which has been similarly transformed into frequency domain. An inverse fast Fourier transform (IFFT) section  23  retransforms the result of this multiplication to time domain, whereby cross-correlation signals pursuant to the respective cyclic shift delays assigned to the user terminals UE 1  to UE 4  can be obtained. In accordance with the uplink or downlink signal reception qualities thus estimated, data rate control is performed. 
         [0019]    When each user&#39;s reception quality is estimated from the cross-correlation signal obtained by the frequency-domain cross-correlation method shown in  FIG. 3A , the following procedure is needed. First, a received signal power is determined for a user in question, based on a received signal in frequency domain, and a received noise component is calculated by subtracting the determined received signal power from the received signal in frequency domain. A reception quality for the user in question is determined by calculating the ratio between the received signal power and the received noise component. For example, assuming that a reference signal exists only in the time period or frequency band in which a user transmits data, it is necessary to estimate a reception quality for each user completely independently of the other users, as shown in  FIG. 3B . 
         [0020]      FIG. 3B  is a block diagram showing an implementation example of the basic configuration shown in  FIG. 3A . To estimate a reception quality for each user completely independently of the other users, after each received signal is transformed to frequency domain by the FFT section  21 , a cross-correlation signal in time domain is obtained for each user through the multiplication processing section  22  and IFFT section  23 . After the signal quality of this cross-correlation signal is improved by a noise reduction section  24 , the signal is transformed again into frequency-domain signals by a FFT section  25 . A received power calculation section  26  calculates a received power Ps on frequency axis, which is then output to each of a subtraction processing section  27  and a reception quality calculation section  28 . 
         [0021]    The subtraction processing section  27  subtracts the received power calculated for the user in question by the received power calculation section  26 , from the received power of the frequency-domain cross-correlation signal obtained by the multiplication processing section  22  through cross-correlation processing, thereby obtaining a noise power Pnz. The subtraction processing section  27  outputs the obtained noise power Pnz to the reception quality calculation section  28 . The reception quality calculation section  28  calculates the ratio between the received power Ps and the noise power Pnz in the frequency domain and estimates a reception quality for the user in question. 
         [0022]    However, according to the circuitry shown in  FIG. 3B , optimal processing cannot be performed for each of control and reference signals in a channel structure like those under consideration in standardization in which control signals, as well as reference signals, are multiplexed by CDM. Moreover, the noise reduction section  24  for performing noise reduction processing on time axis is required for each user to enhance accuracy in the estimation of reception quality. The noise reduction section  24  uses a band-limiting filter and therefore performs complicated processing. 
         [0023]    Furthermore, in the case where different multiplexing schemes are adopted for uplink control signals and data signals, if the estimation of the reception quality of a control signal is performed by using circuitry similar to a data signal reception circuit, then processing suitable for the characteristics of the control signal multiplexing scheme cannot be performed, resulting in the efficiency being degraded. 
       SUMMARY OF THE INVENTION 
       [0024]    Accordingly, an object of the present invention is to provide a reception quality estimation method and device by which reception quality estimation can be performed efficiently. 
         [0025]    According to an aspect of the present invention, a device for estimating reception quality for each of a plurality of user terminals, includes: a correlation extractor for extracting correlation between multiplexed reception reference signals received from the plurality of user terminals and a predetermined reference signal sequence, wherein extracted correlation corresponds to each user terminal; a received power calculator for calculating received signal power from the extracted correlation corresponding to each user terminal; a total received power calculator for calculating total received signal power by summing up the received signal power for all the plurality of user terminals; a noise power estimator for estimating noise power based on the total received signal power and total received power of multiplexed reception reference signals; and a reception quality calculator for calculating the reception quality of each user terminal based on the received signal power and the noise power. 
         [0026]    According to another aspect of the present invention, a method for estimating reception quality for each of a plurality of user terminals, includes: extracting correlation between multiplexed reception reference signals received from the plurality of user terminals and a predetermined reference signal sequence, wherein extracted correlation corresponds to each user terminal; calculating received signal power from the extracted correlation corresponding to each user terminal; calculating total received signal power by summing up the received signal power for all the plurality of user terminals; estimating noise power based on the total received signal power and total received power of multiplexed reception reference signals; and calculating the reception quality of each user terminal based on the received signal power and the noise power. 
         [0027]    According to the present invention, it is possible to efficiently estimate each user&#39;s reception quality from multiplexed reference signals of multiple user terminals. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0028]      FIG. 1  is a schematic diagram to describe cyclic shifts based on a CAZAC sequence. 
           [0029]      FIG. 2A  is a format diagram showing an example of resource allocation in a slot including PUSCH and PUCCH, demodulation reference signals for PUSCH and PUCCH, and a sounding reference signal. 
           [0030]      FIG. 2B  is a block diagram showing a functional configuration of a reference signal sequence generation circuit. 
           [0031]      FIG. 3A  is a block diagram showing a basic configuration of a multi-user channel estimation device. 
           [0032]      FIG. 3B  is a block diagram showing an implementation example of the basic configuration shown in  FIG. 3A . 
           [0033]      FIG. 4A  is a block diagram showing a functional configuration of a reception quality estimation device according to an exemplary embodiment of the present invention. 
           [0034]      FIG. 4B  is a diagram of signal waveforms in time domain showing cross-correlation. 
           [0035]      FIG. 5  is a block diagram showing a specific example of the reception quality estimation device shown in  FIG. 4A . 
           [0036]      FIG. 6A  is a flow chart showing a first example of user-specific delay profile extraction operation in the present exemplary embodiment. 
           [0037]      FIG. 6B  is a flow chart showing a second example of user-specific delay profile extraction operation in the present exemplary embodiment. 
       
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       [0038]    According to an exemplary embodiment of the present invention, reception qualities for multiple users based on a single CAZAC code are collectively estimated, whereby greater efficiency and higher accuracy in reception quality estimation processing can be achieved. Hereinafter, the exemplary embodiment of the present invention will be described in detail by using the accompanying drawings. 
         [0039]      FIG. 4A  is a block diagram showing a functional configuration of a reception quality estimation device according to an exemplary embodiment of the present invention.  FIG. 4B  is a diagram of signal waveforms in time domain showing cross-correlation. According to the present exemplary embodiment, subsequent to a fast Fourier transform (FFT) section  21  which transforms a reference signal received from each user terminal into frequency-domain signals, a processing section is provided correspondingly to each CAZAC code. Here, to avoid complicating the drawing, only the processing section concerning a CAZAC code C 1  is shown. 
         [0040]    Each CAZAC code-specific processing section includes a multiplication processing section  22 , an inverse fast Fourier transform (IFFT) section  23 , and user-specific processing sections  30 -CS 1  to  30 -CSm. As described earlier, received reference signals are transformed by the FFT section  21  into frequency-domain signals, which are multiplied with the CAZAC code C 1  in frequency domain by the multiplication processing section  22 . The result of the multiplication is retransformed into a time-domain signal by the IFFT section  23 , whereby a cross-correlation signal pursuant to the respective cyclic shift delays assigned to user terminals UE 1  to UEm can be obtained as shown in  FIG. 4B . From this cross-correlation signal, the user-specific processing sections  30 -CS 1  to  30 -CSm calculate reception qualities for the multiple user terminals UE 1  to UEm based on the CAZAC code C 1 , on the frequency axis, respectively. Here, m is the number of users that can be multiplexed by cyclic shifts of a CAZAC code in question (in this example, the CAZAC code C 1 ). 
         [0041]    Each of the user-specific processing sections  30 -CS 1  to  30 -CSm includes a delay profile extraction section  31 , a fast Fourier transform (FFT) section  32 , a received power calculation section  33 , and a reception quality calculation section  34 . Only the user-specific processing section  30 -CS 1  concerning the user terminal UE 1  is shown here to avoid complicating the drawing. The configuration and operations of the user-specific processing section  30 -CS 1  concerning the user terminal UE 1  will be described hereinafter as a representative example, since the other user-specific processing sections  30 -CS 2  to  30 -CSm have similar configurations and perform similar operations. 
         [0042]    The delay profile extraction section  31  extracts cross-correlation signal waveforms in the vicinity of (within predetermined bounds from) a cross-correlation power peak corresponding to the user terminal UE 1  and outputs the extracted portion to the FFT section  32  as a delay profile of the user terminal UE 1 . For example, assuming that a CAZAC code made by a one-time cyclic shift of the CAZAC code C 1  is assigned to the user terminal UE 1 , the delay profile of the user terminal UE 1  should appear at a location in time shifted by that cyclic shift delay. Similarly, a delay profile of each of the other user terminals sequentially appears at a location in time shifted by a cyclic shift delay made for the user terminal in question. The extracted portion of the delay profile is determined by predetermined bounds from the cross-correlation power peak corresponding to each user terminal so that the delay spread range for the corresponding user terminal can be included within the extracted portion to more accurately calculate the received signal power for each frequency. 
         [0043]    The thus-obtained delay profile in the vicinity of the peak is transformed to frequency domain by the FFT section  32 . That is, the frequency characteristics of the delay profile for the user terminal UE 1  are output to the received power calculation section  33 . The received power calculation section  33  calculates a received signal power Ps 1  for each frequency by squaring the delay profile in frequency domain and outputs the calculated received signal power Ps 1  to each of the reception quality calculation section  34  and an all-users&#39; received power integration section  35 . In the other user-specific processing sections  30 -CS 2  to  30 -CSm as well, the received power calculation sections  33  calculate received signal powers Ps 2  to Psm for each frequency respectively, which are output to the all-users&#39; received power integration section  35 . Similar operations are performed in the other CAZAC code-specific processing sections. 
         [0044]    In this manner, the all-users&#39; received power integration section  35  receives as input the received signal powers Ps for each frequency, calculated by the respective received power calculation sections  33  of all the user-specific processing sections in every CAZAC code-specific processing section. The all-users&#39; received power integration section  35  sums up the received signal powers Ps and outputs a total received signal power for each frequency to a noise power estimation section  36 . The noise power estimation section  36  calculates a total received power from the frequency-domain signals of the received signals transformed by the FFT section  21 , and subtracts the total received signal power obtained by the all-users&#39; received power integration section  35  from the calculated total received power, thereby estimating a noise power Pnz. The noise power estimation section  36  outputs the estimated noise power Pnz to each of the reception quality calculation sections  34  of all the user-specific processing sections in every CAZAC code-specific processing section. 
         [0045]    The reception quality calculation section  34  calculates the ratio between the received signal power Ps 1 , input from the received power calculation section  33 , and the noise power Pnz, thereby calculating a reception quality for the user terminal UE 1 . Similar processing is also performed in the other user-specific processing sections  30 -CS 2  to  30 -CSm, whereby reception qualities for respective ones of the user terminal UE 2  to UEm are obtained. Moreover, the above-described processing is similarly performed also in the other CAZAC code-specific processing sections. That is, processing similar to the above-described series of processing with respect to the CAZAC code C 1  is performed as many times as the number of multiplexed CAZAC codes while the CAZAC codes are changed. Thus, it is possible to estimate reception qualities for all user terminals. 
         [0046]    Incidentally, the reception quality estimation device shown in  FIG. 4A  can be installed in a base station in a radio communications system such as a mobile communications system. Moreover, not only by the hardware-based circuits, similar functions can also be implemented by executing programs on a program-controlled processor such as CPU. 
       1. SPECIFIC EXAMPLE  
       [0047]      FIG. 5  is a block diagram showing a specific example of the reception quality estimation device shown in  FIG. 4A . Here, it is assumed that a reference extraction section  101  is provided, which extracts reference signals from received signals which are multiplexed as shown in  FIG. 2A , for example, and that a fast Fourier transform (FFT) section  102  transforms the reference signals to frequency domain. These received reference signals on frequency axis are output to CAZAC code-specific processing sections  200 -C 1  to  200 -Cn. Hereinafter, to simplify the description, the configuration and operations of the CAZAC code-specific processing sections will be described by taking the CAZAC code-specific processing section  200 -C 1  as a representative example. 
         [0048]    The CAZAC code-specific processing section  200 -C 1  includes a multiplication processing section  201 , a CAZAC code generation section  202 , an inverse fast Fourier transform (IFFT) section  203 , user-specific processing sections  210 -CS 1  to  210 -CSm, and an all-same-CAZAC-code-users&#39; received power integration section  204 . The all-same-CAZAC-code-users&#39; received power integration section  204  sums up the received signal powers of all users based on the same CAZAC code. The CAZAC code generation section  202  of the CAZAC code-specific processing section  200 -C 1  generates a CAZAC code C 1 . In the other CAZAC code-specific processing sections  200 -C 2  to  200 -Cn as well, the respective CAZAC code generation sections  202  generate CAZAC codes C 2  to Cn, respectively. 
         [0049]    The multiplication processing section  201  performs complex multiplication of the frequency-domain received reference signals input from the FFT section  102  with the frequency-domain reference signal (CAZAC code C 1 ) generated by the CAZAC code generation section  202 , for each frequency. Through this multiplication processing by the multiplication processing section  201 , the received reference signals and the generated reference signal (CAZAC code C 1 ) are multiplied in frequency domain, whereby cross-correlation processing of these signals is performed. The result of this multiplication is transformed to time domain by the IFFT section  203 . Thus, a cross-correlation signal is obtained as shown in  FIG. 4B  that is pursuant to cyclic shift delays respectively assigned to user terminals UE 1  to UEm that are based on the CAZAC code C 1 . 
         [0050]    From this cross-correlation signal, the user-specific processing sections  210 -CS 1  to  210 -Csm respectively calculate reception qualities Qcs 1  to Qcsm for the multiple user terminals UE 1  to UEm multiplexed based on the CAZAC code C 1 . Here, m is the number of users that can be multiplexed by cyclic shifts of a CAZAC code in question (here, the CAZAC code C 1 ). Each of the user-specific processing sections  210 -CS 1  to  210 -Csm includes a peak detection section  211 , a delay profile extraction section  212 , a narrowband fast Fourier transform (FFT) section  213 , a received power calculation section  214 , and a reception quality calculation section  215 . Note that the peak detection section  211  and delay profile extraction section  212  correspond to the delay profile extraction section  31  in  FIG. 4A . Here, to avoid complicating the drawing, only the user-specific processing section  210 -CS 1  concerning the user terminal UE 1  is shown. Hereinafter, the configuration and operations of the user-specific processing section  210 -CS 1  concerning the user terminal UE 1  will be described as a representative example, since the other user-specific processing sections  210 -CS 2  to  210 -CSm have similar configurations and perform similar operations. 
         [0051]    First, the peak detection section  211  detects the location of a peak corresponding to the reception timing of the user terminal UE 1 , from the signal on the time axis input from the IFFT section  203  (see  FIG. 4B ). The delay profile extraction section  212  detects a delay profile of the user terminal UE 1 , according to the detected peak location. Specifically, the delay profile extraction section  212  extracts cross-correlation signal waveforms within a predetermined range from the detected peak location corresponding to the user terminal UE 1  and outputs the extracted portion to the narrowband FFT section  213  as a delay profile of the user terminal UE 1 . 
         [0052]    The narrowband FFT section  213  transforms the delay profile of the user terminal UE 1  again to frequency domain and outputs the frequency-domain delay profile to the received power calculation section  214 . Incidentally, “narrowband” of the narrowband FFT section  213  means that the number of points for Fourier transform is smaller than those of the ordinary FFT section  102  and IFFT section  203 . The number of points is a number obtained by dividing the number of points for ordinary FFT by the number of users that can be multiplexed based on the same CAZAC code. The received power calculation section  214  squares the thus-obtained frequency-domain delay profile for each frequency, thereby calculating the value of received signal power at this point in time for the user terminal UE 1 , for each frequency. 
         [0053]    The received signal powers of all user terminals UE 1  to UEm based on the CAZAC code C 1 , obtained in this manner, are summed up by the all-same-CAZAC-code-users&#39; received power integration section  204 , whereby a total users&#39; received signal power Psc 1  with respect to the CAZAC code C 1  is calculated and output to an all-users&#39; received power integration section  222 . Similarly, with respect to the other CAZAC codes C 2  to Cn as well, the received signal powers of multiplexed user terminals are summed up by the respective all-same-CAZAC-code-users&#39; received power integration sections  204 , and obtained total users&#39; received signal powers Psc 2  to Pscn with respect to the CAZAC codes C 2  to Cn are output to the all-users&#39; received power integration section  222 . The all-users&#39; received power integration section  222  sums up the total users&#39; received signal powers Psc 1  to Pscn with respect to the CAZAC codes C 1  to Cn, whereby an all-users&#39; signal power sum total value Ps is calculated and output to a subtraction processing section  221 . 
         [0054]    On the other hand, a total received power estimation section  220  calculates a total received power value Ptotal, which is the total power of all received signals, from the frequency-domain received reference signals input from the FFT section  102 . The subtraction processing section  221  subtracts the all-users&#39; signal power sum total value Ps from the total received power value Ptotal input from the total received power estimation section  220 , thereby calculating a noise power Pnz for each frequency. The subtraction processing section  221  outputs the calculated noise power Pnz to all the reception quality calculation sections  215  in every CAZAC code-specific processing section. 
         [0055]    Each reception quality calculation section  215  calculates the ratio between the received signal power Ps of its corresponding user terminal UE and the noise power Pnz input from the subtraction processing section  221 . Thus, the signal to noise ratios, which are the reception qualities for the respective user terminals, can be obtained as the reception qualities Qcs 1  to Qcsm for the user terminals UE 1  to UEm, respectively. 
       2. EXTRACTION OF DELAY PROFILE  
       [0056]    Next, the operations of the peak detection section  211  and delay profile extraction section  212  (corresponding to the delay profile extraction section  31  in  FIG. 4A ) will be described in more detail. The operations of these sections can also be implemented by executing a program on CPU. 
       2.1) First Example 
       [0057]      FIG. 6A  is a flow chart showing a first example of the user-specific delay profile extraction operation in the present exemplary embodiment. First, the peak detection section  211  extracts a temporal portion where a delay profile of its corresponding user terminal exists (Step S 301 ). This temporal portion can be uniquely determined based on the cyclic shift amount because users are multiplexed by cyclic shifts of a CAZAC code. For example, assuming that this peak detection section  211  belongs to the user-specific processing section for the user terminal UE 1 , this extracted temporal portion is the temporal portion corresponding to the user terminal UE 1  in  FIG. 4B . 
         [0058]    Subsequently, the peak detection section  211  squares the cross-correlation signal in the extracted temporal portion, thereby converting the signal into power values (Step S 302 ). The peak detection section  211  then detects the largest value (peak power value) of the power values, as well as the location of this value (peak location) (Step S 303 ). 
         [0059]    The delay profile extraction section  212  samples a fixed number of times the cross-correlation signal only in the vicinity of the detected peak location (within predetermined bounds from the peak location as the median), thereby generating a delay profile of the user terminal in question (Step S 304 ). 
       2.2) Second Example 
       [0060]      FIG. 6B  is a flow chart showing a second example of the user-specific delay profile extraction operation in the present exemplary embodiment. First, the peak detection section  211  detects a peak power value and a peak location as in the above-described Steps S 301  to S 303  (Steps S 401  to S 403 ). In the second example, the generation of a delay profile is controlled depending on whether or not the peak power value is equal to or greater than a predetermined level. 
         [0061]    First, the delay profile extraction section  212  determines whether or not the detected peak power value is equal to or greater than a predetermined threshold value (Step S 404 ). When the peak power value is equal to or greater than the predetermined threshold value (Step S 404 : Yes), the delay profile extraction section  212  samples a fixed number of times the cross-correlation signal only in the vicinity of the detected peak location (within the predetermined region from the peak location as the median) and thereby generates a delay profile of its corresponding user terminal (Step S 405 ). On the other hand, when the peak power value is smaller then the predetermined threshold value (Step S 404 : No), the delay profile extraction section  212  generates data in which all values are zero, as a delay profile of the user terminal in question (Step S 406 ). Thus, the reception quality for the user terminal in question is made to be zero, which is the lowest value. 
         [0062]    As described above, when the peak power value is equal to or greater than the threshold value, it is determined that the reception quality at the user terminal in question is worth subjecting to data rate control, and the same processing as shown in  FIG. 6A  is performed. However, when the peak power value is smaller than the threshold value, it is determined that the reception quality at the user terminal in question is not enough to be subjected to data rate control, and more bands are allocated to a user exhibiting better reception quality. Thus, it is possible to enhance the total throughput. 
       3. ASPECTS OF THE PRESENT INVENTION  
       [0063]    As described above, a reception quality estimation device according to the present invention extracts a cross-correlation profile of each of multiple user terminals and calculates a received signal power of the cross-correlation profile of each user terminal in frequency domain. Further, the reception quality estimation device estimates a noise power based on a total received signal power, which is obtained by summing up the received signal powers of all the multiple user terminals, and on a total received power, which is the total power of all received reference signals in frequency domain. Based on the received signal power of each user terminal and the noise power, the reception quality estimation device estimates a reception quality for each user terminal. 
         [0064]    The extraction of a cross-correlation is performed as follows. A cross-correlation is transformed to time domain, from which the location in time of a peak corresponding to each user terminal is detected. For each user terminal, a cross-correlation within a predetermined region from the peak location in time is extracted as a cross-correlation profile, which is transformed to frequency domain. Thereby, individual cross-correlation profiles can be generated. 
         [0065]    Preferably, a cross-correlation peak value at the peak location in time is further detected, and when the cross-correlation peak value is equal to or greater than a predetermined threshold value, a cross-correlation profile is effectively extracted. 
         [0066]    According to the present invention, it is possible to perform reception quality estimation exploiting the property of CAZAC codes. For example, in a radio communications system in which data rate control is performed depending on the reception quality of an uplink or downlink signal, a signal reception quality for each user terminal can be estimated with efficiency from reference signals of multiple user terminals multiplexed. High-speed, lightweight operation can be accomplished because noise reduction processing, such as filtering on the time axis for each user, is not required. Moreover, the amount of processing can be further reduced by using a narrowband fast Fourier transformer (FFT) to transform a cross-correlation profile into frequency-domain signals. Consequently, it is possible to efficiently calculate the frequency property of reception quality for each user terminal. 
         [0067]    According to a specific example of the present invention, as shown in  FIG. 5 , the multiplication processing section  201  multiplies a frequency-domain received signal obtained by the FFT section  102 , by a frequency-domain reference signal corresponding to a multiplexed CAZAC code. For each of users multiplexed by cyclic shifts of the same CAZAC code, the peak detection section  211  and delay profile extraction section  212  extracts a delay profile of each user from the time-domain signal obtained by the IFFT section  203 , and the narrowband FFT section  213  calculates the frequency characteristics of the received power. Further, the total received power estimation section  220  estimates a total received power, which is the total power of all received signals on the frequency axis, and the subtraction processing section  221  calculates the frequency characteristics of a noise signal by subtracting an all-users&#39; received signal power from the total received power. For each user, the reception quality calculation section  215  calculates the ratio between the received signal power of the user in question and the noise signal on the frequency axis, thereby estimating frequency-domain reception quality for the user in question. If the above-described series of processing is performed as many times as the number of the CAZAC codes multiplexed, it is possible to estimate reception quality for all users. 
         [0068]    Hence, according to the present invention, it is possible to efficiently calculate the frequency characteristics of reception quality for each user by using a narrowband FFT, which performs a small amount of processing, without requiring noise reduction processing such as filtering on the time axis for each user. 
         [0069]    The present invention can be applied to radio communications system in which data rate control is performed depending on the reception quality of an uplink or downlink signal. 
         [0070]    The present invention may be embodied in other specific forms without departing from the spirit or essential characteristics thereof. The above-described exemplary embodiment and examples are therefore to be considered in all respects as illustrative and not restrictive, the scope of the invention being indicated by the appended claims rather than by the foregoing description, and all changes which come within the meaning and range of equivalency of the claims are therefore intended to be embraced therein.