Abstract:
The present invention is about a method and apparatus for direct sequence spread spectrum (DS/SS) system receivers that employ a channel estimator for coherent demodulation. The proposed DS/SS receiver is composed of the channel estimation controller (CEC), the channel parameter estimator (CPE), the channel estimation filter (CEF) module and the demodulator module. The proposed scheme estimates the channel environment when the pilot signal is transmitted in parallel with the message signal for coherent detection. The internal parameters of the CPE and the CEF are determined by the CEC according to the operating conditions. The CPE adaptively controls the bandwidth of the CEF by classifying the channel type from the received pilot signal. Thus, the proposed scheme can obtain improved channel information, providing the enhancement of the receiver performance. This enhancement can provide the improvement of the link performance, the service time and/or the user capacity.

Description:
BACKGROUND OF THE INVENTION 
   The transmitter of a direct sequence spread spectrum (DS/SS) system spreads narrowband message signals into wideband signals using the spreading sequences. The receiver detects the desired signal by multiplying the received signal with the spreading sequences synchronized to that of the transmitter is disclosed in J. G. Proakis,  Digital communications , McGraw-Hill, 3rd edition, 1995. To receive the signal, coherent receiver scheme needs the phase information of the received signal, providing better reception performance than noncoherent one at the expense of increased complexity. In particular, coherent detection in a rake receiver requires channel information including the amplitude and the phase of the channel. The channel information can be estimated using a pilot signal transmitted with the message signal. An example is disclosed in F. Ling, “Coherent detection with reference symbol based channel estimation for direct sequence CDMA uplink communications,”  IEEE Proc. VTC  &#39;93, pp. 400–403, May 1993. 
   The accuracy of channel estimation can be improved by reducing the noise effect contained in the obtained channel information, which is usually achieved by employing a lowpass filter, called the channel estimation filter (CEF). The performance of channel estimation is significantly affected by the CEF and the channel condition, including the maximum Doppler frequency and fading characteristics. In general, commercial CDMA receivers employ conventional CEF with fixed parameters (e.g., the cut-off bandwidth). 
   A DS/SS receiver employing an adaptive channel estimator was proposed in Kitade et al.,  Spread spectrum receiving apparatus , U.S. Pat. No. 6,134,262, 2000, where the CEF is changed based on the maximum Doppler frequency. The maximum Doppler frequency was estimated by measuring how many times the received signal power exceeds a predetermined threshold. However, the scheme in said Kitade et al.,  Spread spectrum receiving apparatus , U.S. Pat. No. 6,134,262, 2000, may not be practical when the transmitted signal is fast power controlled for compensating fast fading as well as the near-far effect, because the power of the received signal cannot fully reflect the fading statistics of the channel. 
   BRIEF SUMMARY OF THE INVENTION 
   The present invention is about a method and apparatus for reception of DS/SS signals that improves the receiver performance by enhancing the accuracy of the estimated channel information using an adaptive channel estimator. 
   The channel estimator may behave differently depending upon the channel environment. The parameters of a conventional CEF are designed for a fixed, particular channel condition. As a result, the performance of the channel estimator employing a CEF with fixed parameters can be degraded when the channel condition becomes different from one assumed for the design of the CEF. It can be expected that improved channel estimation can be obtained if the parameters of the CEF are adaptively adjusted according to the channel environment. The operating condition can be estimated using the received pilot signal. Although the scheme in said Kitade et al.,  Spread spectrum receiving apparatus , U.S. Pat. No. 6,134,262, 2000 considers only the maximum Doppler frequency, the present invention considers the fading statistics, the number of multipaths and the received signal power as well as the maximum Doppler frequency. This enables the present invention to provide improved channel estimation performance regardless of the employment of fast power control. As a result, the present invention can improve the system performance, making it possible to improve the link performance, the user capacity and/or the service time. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is the l-th finger of a proposed DS/SS receiver using an adaptive channel stimator; 
       FIG. 2  is the channel estimation controller (CEC) of  FIG. 1 ; 
       FIG. 3  represents design procedure that the CEC performs; 
       FIG. 4  depicts an example of parameter decision of the CEC; 
       FIG. 5  is the channel parameter estimator (CPE) of  FIG. 1 ; 
       FIG. 6  represents the structure of correlator bank in the CPE; 
       FIG. 7  is the channel estimation filter module of  FIG. 1 ; 
       FIG. 8  represents a CEF employing a moving average (MA) FIR filter; 
       FIG. 9  represents a CEF employing a general FIR filter; 
       FIG. 10  represents a CEF employing a general IIR filter; and 
       FIG. 11  represents a CEF employing a single-pole IIR filter. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 1  depicts the ι-th finger of the proposed DS/SS receiver employing an adaptive channel estimator. The proposed DS/SS receiver is composed of the channel estimation controller (CEC)  102 , the channel parameter estimator (CPE)  101 , the channel estimation filter (CEF) module  103  and the demodulator module. The CEC  102  determines the internal parameters of the CPE  101  and the CEF module  103  considering the characteristics of operating condition. The CPE  101  classifies the channel type using the received pilot signal and selects a CEF appropriate to the estimated channel type. The received signal is despread using a spreading sequence synchronized with that of transmitter at each finger and then combined using the channel gain obtained by the CEF in the demodulator module. 
   The received signal r(t+d 1 T c ) of the ι-th path is despread by the multiplier  108  that multiplies an appropriate PN sequence synchronized with that of the transmitter. The multiplier  106  separates the pilot signal from the data signal by multiplying the input signal with orthogonal pilot sequence. The accumulator  104  accumulates the pilot signal for an amount of samples equal to the spreading factor to obtain the pilot symbol ĥ l [k]. Similarly, data symbol ŷ l [k] is generated using the orthogonal data sequence. 
   The channel estimator  100  provides the channel information for combining the rake receiver outputs. The CEC  102  generates the control signals that initialize the internal parameters of the CPE  101  and the CEF module  103  considering the channel environment and operating condition. The CPE  101  provides the CEF module  103  with the channel parameters using the received pilot signals. The parameters of the CEF are determined based on the output of the CPE  101 . The CEF can be implemented in the form of a finite impulse response (FIR) filter or infinite impulse response (IIR) filter. For example, if a moving average (MA) type FIR filter is used as the CEF, as shown in  FIG. 8 , the number of taps can be determined based on the output of CPE  101 . When a general type FIR filter is used as the CEF as shown in  FIG. 9 , the coefficients and the tap size of the CEF is selected among the pre-designed FIR filters considering the output of the CPE  101 . If an IIR filter is used as the CEF as shown in  FIG. 10 , one of pre-designed IIF filters is selected based on the output of CPE  101 . As a simple case, a single-pole IIR filter can be used as the CEF as shown in  FIG. 11 , where the forgetting factor is determined based on the output of CPE  101 . 
   The channel information can be obtained by filtering the pilot symbols {tilde over (h)} l [k] using the CEF. The data symbols ŷ l [k] pass through the delay element  109  to compensate the delay due to the channel estimation process. The delayed data symbol is multiplied by the conjugate of the estimated channel gain using the multiplier  110 . Finally, all the received data signals from each finger of the rake receiver are combined for decision. 
   The operation of the proposed adaptive channel estimator is as follows. The CEC  102  in  FIG. 1  is redrawn in  FIG. 2 . The CEC  102  takes norminal operating condition  5  parameters required by the base station or mobile station to provide appropriate services. These parameters for the l-th finger include the minimum and the maximum tap size, N l,min  and N l,max , the minimum and the maximum values of the ratio of total received power to the receiver power of the l-th path, A l,min  and A l,max , the minimum and the maximum values of the channel spectrum index, χ l,min  and χ l,max , the minimum and the maximum values of the Ricean factor, K l,min  and K l,max , the minimum and the maximum values of the incident angle of the direct ray, |θ l | min  and |θ l | max , the minimum and the maximum values of the pilot to data signal power ratio, β min  and β max  and the minimum and the maximum values of the signal to noise power ratio per bit, γ b min  and γ b max . In addition, the CEC  102  requires the information on the code rate R, the threshold value η, the noise rejection ratio κ p  of the prefilter  501  which means the noise suppression factor equal to the input noise to output noise power ratio of the prefilter, and upper bound and lower bound of the tap margin, ε L  and ε U . This margin is related to the decision of the tap size of the CEF and the correlation interval of the correlator and 0&lt;ε L ≦ε U &lt;1, in general. Based on these parameters, the CEC  102  determines the number G l  of correlators  502 , the set m l  of the delayed symbols m l,i  of the CPE  101 , and the set N l  of the CEF tap size N l,i . 
   The design procedure of the CPE  101  is depicted in  FIG. 3 , where the internal variables ζ l,min  and ζ l,max  used for steps  302  and  303  can be calculated in step  301  using Eq. [1] and [2]. 
                 ζ     1   ,   min       =         y   min     -         y   min   2     -     4   ⁢     x   min     ⁢     z   min               2   ⁢     x   min           ⁢     
     ⁢   where   ⁢     
     ⁢       x   min     =         K     l   ,   max       ⁢     cos   4       |     θ   l     ⁢     |   min     ⁢       /   24     +     1   /   64           ⁢     
     ⁢       y   min     =         K     l   ,   max       ⁢     cos   2       |     θ   l     ⁢     |   min     ⁢       /   2     +     1   /   4           ⁢     
     ⁢       z   min     =         (     1   -   η     )     ⁢     (     1   +     K     l   ,   max         )       -       η   ⁢           ⁢       A     l   ,   max       ⁡     (     1   +     K     l   ,   max         )           R   ⁢           ⁢     β   min   2     ⁢     κ   p     ⁢     γ     b   ⁢           ⁢   min                       Eq   .           ⁢     [   1   ]               
 
where
 
Here, the subscripts max and min respectively denote the maximum and minimum values of the parameter. 
                 ζ     1   ,   min       =         y   max     -         y   max   2     -     4   ⁢     x   max     ⁢     z   max               2   ⁢     x   max           ⁢     
     ⁢   where   ⁢     
     ⁢       x   max     =         K     l   ,   min       ⁢     cos   4       |     θ   l     ⁢     |   max     ⁢       /   24     +     1   /   120           ⁢     
     ⁢       y   max     =         K     l   ,   min       ⁢     cos   2       |     θ   l     ⁢     |   max     ⁢       /   2     +     1   /   6           ⁢     
     ⁢       z   max     =         (     1   -   η     )     ⁢     (     1   +     K     l   ,   min         )       -       η   ⁢           ⁢       A     l   ,   min       ⁡     (     1   +     K     l   ,   min         )           R   ⁢           ⁢     β   max   2     ⁢     κ   p     ⁢     γ     b   ⁢           ⁢   max                       Eq   .           ⁢     [   2   ]               
 
where
 
   The tap size of the first CEF, N l,1 , is set to the minimum tap size N l,min . Then, using the lower bound margin ε L , the amount of delay m l,1  of the first correlator  502  can be determined in step  302  using Eq. [3] with i=1 such that the point A in  FIG. 4  is represented as (m l,1 ,(1+ε L ) −1 N l,1 ). 
               m     l   ,   i       =     0.5   ⁡     [               ⁢         (         (     1   +     ε   L       )       -   1       ⁢     N     l   ,   i         )     5     ⁢   R   ⁢           ⁢     γ     b   ⁢           ⁢   max       ⁢     ζ     l   ,   max     2     ⁢       β   max   2     ⁡     (         K     l   ,   min       ⁢     cos   4       |     θ   l     ⁢     |   max     ⁢       /   9     +     1   /     χ     l   ,   max             )               A     l   ,   min       ⁡     (     1   +     K     l   ,   min         )         ]               Eq   .           ⁢     [   3   ]               
 
   Using an upper boundary margin ε U  and m l,1 , the tap size N l,2  of the second CEF in the l-th finger can be calculated such that (m l,1 ,(1+ε U )N l,2 ) is on the upper boundary (say, the point B in  FIG. 4  for i=1). Thus, N l,2  can be determined in step 303 using Eq. [4] with i=1, 
               N     l   ,     i   +   1         =       1     1   +     ε   U         ⁢       (       16   ⁢     A     l   ,   max       ⁢       m     l   ,   i     4     ⁡     (     1   +     K     l   ,   max         )             Rγ     b   ⁢           ⁢   min       ⁢     β   min   2     ⁢       ζ     l   ,   min     2     ⁡     (         K     l   ,   max       ⁢     cos   4       |     θ   l     ⁢     |   min     ⁢       /   9     +     1   /     χ     l   ,   min             )           )       1   /   5                 Eq   .           ⁢     [   4   ]               
 
   Similarly, m l,2  can be calculated using N l,2  and Eq.[3] with i=2. In this way, m l,i , i=1, 2, . . . , G l , and N l,i , i=1, 2, . . . , G l +1, can be calculated iteratively until the tap size becomes larger than or equal to the predetermined maximum value N l,max . When N l,i+1 ≧N l,max , G l  and N l,i+1  are set to a value of i and N l,max , respectively. 
     FIG. 4  illustrates a design example of the parameters for the CPE  101 , where the upper bound is determined by the values of A l,min , χX l,min , K l,max , |θ l | min , β min  and γ bmin  and the lower bound by the values of A l,max , χ l,min , K l,min , |θ l | max , β max  and γ bmax . For example, when N l,min =11, N l,max =150, A l,min =1, A l,max =6, χ l,min =24, χ l,max =45, K l,min =0, K l,max =∞, |θ l | min =0, |θ l | max =90°, β min =β max =¼, γ bmin =3 dB, γ bmax =10 dB, ε L =ε U =0.3, R=½, κ p =12 and η=0.3, the value of G l  is determined to 5. Then, the values of m l,i  and N l,i , i=1, 2, . . . , 5 obtained by the procedure shown in  FIG. 3 , are m l ={12, 24, 48, 96, 192}, N l ={11, 17, 30, 53, 91, 145}. 
     FIG. 5  depicts the structure of the CPE  101  of  FIG. 1 . The received pilot symbol {tilde over (h)} l [k] is first lowpass filtered by the prefilter  501  to reduce the noise. A conventional lowpass filter including an MA FIR filter or a simple IIR filter can be used as the prefilter  501 . The prefiltered pilot symbol {overscore (h)} l [k] is input to the correlator bank  502  which comprises G l  correlators, where the number G l  is determined by the CEC  102 . 
   The detail block diagram of the correlator bank 502 is depicted in  FIG. 6 . In the i-th correlator, the filtered pilot symbol {overscore (h)} l [k] is correlated with the m l,i  symbol delayed pilot signal {overscore (h)} l [k−m l,i ] using the accumulator  602  for an interval of J symbols. The output of the correlator is normalized by the module  603  by using Eq. [5]. Since the values of m l,i  are determined in the CEC  102  such that m l,i &lt;m l,i+1 , i=1, 2, . . . , G l −1, the correlator output w l,i  fast decreases as i increases. 
               w     l   ,   i       ≡         ∑   J     ⁢     Re   ⁢     {             h   _     l     *     ⁡     [   n   ]       ⁢         h   _     l     ⁡     [     n   -     m   i       ]         }             ∑   J     ⁢     |         h   _     l     ⁡     [   n   ]       ⁢     |   2                   Eq   .           ⁢     [   5   ]               
 
   The outputs of correlators, w l,1 , w l,2 , . . . , and w l,G     l   , are sequentially compared with a given threshold η by the comparator  503 . The comparator  503  finds out which correlator output is less than η for the first time by increasing the correlator index number i from i=1 to G l . If the j-th correlator output becomes less than η for the first time, α l  is set to j, 1≦j≦G l . When there is no correlator output less than η, α l  is set to a value of G l +1. 
     FIG. 7  depicts the structure of the CEF module  103 . The tap size selector  701  determines the tap size N l,α     l    according to the output α l  of the CPE  101  among the values of N l,i , i=1, 2, . . . , G l +1. The structure of the CEF is depicted in  FIG. 8  when an MA FIR filter is used as the CEF. A maximum number, N l,max , of the pilot symbols {tilde over (h)} l [k], are sequentially stored in the register  801 . If N l,α     l    is chosen by the tap size selector  701 , the contents of the leftmost N l,α     l    registers in the register  801  are added by the adder  802  and then averaged by the averaging module  803 , yielding the channel information as Eq. [6]. 
                   h   ^     l     ⁡     [     k   -     ⌊       N     l   ,     α   l         /   2     ⌋       ]       =       1     N     l   ,     α   l           ⁢       ∑     i   =   0         N     l   ,     α   l         -   1       ⁢         h   ~     l     ⁡     [     k   -   i     ]                   Eq   .           ⁢     [   6   ]                 
where └x┘ denotes an integer k such that k≦x≦k+1. In this case, the output of the delay element  109  is delayed by └N l,α     l   /2┘ symbols.
 
   The structure of the CEF is depicted in  FIG. 9  when general FIR filters are used as the CEF. FIR filters are designed such that they have a cut-off frequency similar to that of an N l,α     l   -tap MA FIR filter, α l =1, 2, . . . and G l +1. The coefficients {α l } are selected among (G l +1) coefficient sets of designed FIR filters. The number N′ l,max  is the largest tap size of the designed FIR filter corresponding to an equivalent N l,max -tap MA CEF. Finally, the channel information is obtained by adding the received pilot symbols weighted by coefficients {α l } as Eq. [7]. 
                   h   ^     l     ⁡     [     k   -     ⌊       N     l   ,     α   l       ′     /   2     ⌋       ]       =       ∑     i   =   0         N     l   ,     α   l       ′     -   1       ⁢       α     l   ,   i       ⁢         h   ~     l     ⁡     [     k   -   i     ]                   Eq   .           ⁢     [   7   ]               
 
When the tap size of the designed FIR filter corresponding to an N l,α     l   -tap MA FIR filter is N′ l,α     l   , the output of the delay element  109  is delayed by └N′ lα     l   /2┘ symbols.
 
   The structure of the CEF is depicted in  FIG. 10  when IIR filters are used as the CEF. These IIR filters are designed such that they have a cut-off frequency similar to that of an N l,α     l   -tap MA filter, α l =1, 2, . . . and G l +1. The coefficients {α l } and {β l } are selected among the coefficient sets of the designed IIR filters. The channel information is obtained by passing the despread pilot symbol into the selected IIR filter as Eq. [8]. The output of the delay element  109  is delayed by N l,IIR  symbols considering the group delay of the selected IIF CEF. 
                   h   ^     l     ⁡     [     k   -     N     l   ,   IIR         ]       =         ∑     i   =   0         N     l   ,     α   l       ′     -   1       ⁢       α     l   ,   i       ⁢         h   ~     l     ⁡     [     k   -   i     ]           +       ∑     i   =   0         N     l   ,     α   l       ′     -   1       ⁢       β     l   ,   i       ⁢         h   ^     l     ⁡     [     k   -     N     l   ,   IIR       -   i     ]                     Eq   .           ⁢     [   8   ]               
 
   The structure of the CEF is depicted in  FIG. 11  when a single-pole IIR filter is used as the CEF. The forgetting factor α l,IIR  is set to (N l,α     l   /2−1)/(N l,α     l   /2) when N l,α     l    is chosen by the tap size selector  701 . The channel information is obtained using Eq. [9]. The output of the delay element  109  is delayed by └N l,α     l   /2┘ symbols. 
                 h   ^     l     ⁡     [         k   -     ⌊       N     l   ,     α   l         /   2     ⌋       |     =         (     1   -     α     l   ,   IIR         )     ⁢         h   ~     l     ⁡     [   k   ]         +       α     l   ,   IIR       ⁢       h   ^     l       -     ⌊       N     l   ,     α   l         /   2     ⌋     -   1       ]             Eq. [9]