Abstract:
The invention concerns a time domain reflectometry method for estimating properties of a transmission channel, for instance a channel for transmitting electric or acoustic signals. This method includes the steps of generating, at one end of the channel, a plurality of pulses ( 40, 42, 44 ) covering differency frequency bands, and processing the echoes provided by these pulses at the same end of the channel. The frequency bands of the generated pulses are preferably overlapping. The invention is particularly applicable to the testing of ADSL services.

Description:
BACKGROUND OF THE INVENTION 
   The invention relates to a method and apparatus for determining properties of a transmission line or channel, for instance a channel for transmitting electric or acoustic signals. It concerns also a time domain reflectometry method and an equipment for implementing this method. 
   In the field of telecommunications, the density of transmitted information increases regularly. This increase is not always compatible with the existing equipment, more particularly with the existing transmission lines. In order to cope with this problem, several technologies are known, for instance ADSL which means “Asymmetric Digital Subscriber Line” (or other DSL services such as HDSL and VDSL). This technology provides the possibility to transmit, with ordinary telephone lines, high data rate from the network (a central office) to the subscriber and lower data rates from the subscriber to the network. 
   ADSL is adapted for distances, between a central station or office and the subscriber, which are comprised between 1.5 and 6 km. HDSL is adapted for distances greater than 6 km, and VDSL is adapted for distances comprised between 0.3 and 1.5 km. 
   In order to be able to comply with ADSL services, the telephone line must satisfy standards of quality which are not necessarily fulfilled by all telephone lines. 
   SUMMARY OF THE INVENTION 
   Before implementing ADSL services, it is therefore necessary to evaluate a priori the line quality at the lowest cost as possible and with the highest accuracy as possible. The present invention relates to a method and an apparatus which determine, from a central office, the transmission characteristics of a transmission line with a high signal to noise ratio and a high accuracy. 
   The invention thus contributes to the line qualification. 
   Up to now an a priori line qualification requires to make tests at both ends of the line. This means that one operator must be present at the central office and another operator at the subscriber&#39;s location. Although accurate, this a priori line testing is expensive. 
   The method according to the invention is characterized in that, for testing a line, use is made of time domain reflectometry wherein, at one end of the line, a plurality of pulses covering different bandwidths are transmitted in the line and the corresponding echoes are processed, more particularly filtered and reconstructed, at the same end, the echoes providing information about the whole length of the line. 
   Up to now, time domain reflectometry has been used only for detecting the first defect on a line. Time domain reflectometry is known to use only one pulse of narrow or wide frequency band at one time and the transmitted pulse provides only one echo. With the wide band “multifocal” (multi echoes) time domain reflectometry according to the invention, several echoes are provided which give information about the entire length of the line. In fact, with the invention, which makes use of pulses at different bandwidths, it is possible to optimize both the time accuracy and, with an appropriate filtering, the signal to noise ratio along the whole line, at any distance from the central office. 
   The plurality of pulses, called here “multifocality”, allows to partially compensate for limited dynamics of the A/D (analog to digital) converters. In fact, with only one wide band pulse, it would be necessary to use a pulse of very high amplitude in comparison to the A/D converter ability. 
   In a preferred embodiment, the pulses are analytical complex pulses and contribute to a high signal to noise ratio. 
   Preferably the bandwidths of the transmitted pulses are overlapping and are such that the reconstructed echoes present a flat spectrum, i.e. a perfect line. In other words, without defects and without attenuation, the response has practically a constant amplitude over the whole useful frequency spectrum. In this way, the method provides directly the reflectometric impulse response of a line, for instance of a copper pair, generally a twisted pair. 
   As the pulses cover different bandwidths, in order to obtain this flat spectrum, it is necessary that, after reflection and processing, the modulus of the sum of the complex responses of the overlapping regions equals the modulus (for instance one) in the non-overlapping region. 
   In a preferred embodiment, the spectrum amplitude on one side of a band-width is decreasing according to a sine function and the overlapping part of the neighboring bandwidth is increasing according to a cosine function and the detected signal is the sum of squares of all components. 
   With this embodiment, after reflection, the first end of the first bandwidth will vary as sin 2  and the overlapping adjoining beginning of the following bandwidth will vary as a cosin 2 . Therefore, in the overlapping region: sin 2 +cos 2 =1. 
   With this embodiment, it is also possible to provide a given gain (or attenuation) to each bandwidth in order that, for each of these bandwidths, the pulse be compatible with the power limitations imposed by standards. For instance, ADSL requests that the power be limited to −40 dbm/hertz (10 −4  mwatt/hertz) and VDSL requests a limitation to −60 dbm/hertz (10 −6  mwatt/hertz) for frequencies higher than 1 MHz. 
   Therefore, in order to obtain a flat spectrum, for each bandwidth, the transmitted pulse has a given amplification (or attenuation) and the received echo is provided with a complementary attenuation (or amplification). 
   The echoes provide in the time domain, and/or in the frequency domain, an information about the properties, generally the defects, of the line. Moreover, the positions in time of the echoes represent the locations of the defects. 
   In an embodiment, the frequency bandwidth and the amplitude of the low frequency pulses are selected according to the line attenuation and its compliancy in terms of egress. An egress compliant line is a line which does not disturb neighbouring services. In other word means are provided for selecting the frequency bandwidth and the amplitude of the low frequency pulses. 
   The time domain reflectometry method according to the invention is not limited to the estimation of the attenuation of telephone lines. More generally, this method may be used for estimating transmission channels which are dispersive, and subject to attenuation and noise. 
   These channels are not necessarily channels for transmitting electric signals; the signals may be of a different nature, for instance acoustic signals. The method provides the reflectometric impulse response with a good accuracy and a good signal-to-noise ratio at any distance. 
   In brief, the invention concerns a time domain reflectometry method for estimating properties of a transmission channel, for instance a channel for transmitting electric or acoustic signals, which is characterized in that it comprises the steps of generating, at one end of the channel, a plurality of pulses covering different frequency bands, and of processing the echoes provided by these pulses at the same end of the channel. The invention therefore determines a priori the reflectometric impulse response of this transmission line since it is to be noted that a spectrum over a wide bandwidth is equivalent to a Dirac function. 
   The invention relates also to a method for testing the properties, such as the attenuation, of telephone lines comprising copper pairs, for instance twisted pairs, between a central office and a subscriber, which is characterized in that it makes use of the time domain reflectometry. 
   The invention provides an apparatus for testing the properties, such as the attenuation, of telephone lines comprising copper pairs, for instance twisted pairs, between a central office and a subscriber, which is characterized in that it comprises time domain reflectometry means. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     Other features and advantages of the invention will appear with the following description of certain of its embodiments, this description being made in connection with the following drawings, wherein: 
       FIG. 1  represents the application of a method according to the invention to the estimation of defects of a telephone line for ADSL services, 
       FIG. 2  is a schematic diagram showing an equipment according to the invention, this diagram representing also the operation of the equipment, i.e. the method according to the invention, 
       FIGS. 3   a  and  3   b  show an example of pulses generated by the equipment represented on  FIG. 2 , and 
       FIGS. 4   a  through  4   f ,  FIGS. 5   a  through  5   f ,  FIGS. 6   a  through  6   f , and  FIGS. 7 and 8  are diagrams showing signals at different locations on the receiving side of the equipment represented on FIG.  2 . 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   The time domain reflectometry method according to the invention was developed in order to test copper lines  10  between a central office  12  of a telephone operator and subscribers  14 . The goal of the test is to check whether the copper pair  10  is able to comply with ADSL services, i.e. whether its attenuation is inferior to a maximum attenuation imposed by standards. Moreover, this method provides the possibility to localize defects of the line  10 . 
   In the schematic example shown on  FIG. 1 , a part  16  of the line  10  which is relatively dose to the central office or station  12  is subject to humidity; after this defect  16 , in the direction from the office  12  to the subscriber  14 , the line presents a defect  18  corresponding to a bridge tap, i.e. a derivation. 
   In the preferred embodiment of the method according to the invention, a set of pulses are simultaneously or sequentially generated (sequentially in the example) at the central station  12 , each pulse covering a given bandwidth and the bandwidths of all the pulses are overlapping. The whole bandwidth formed by these pulses is wide, for instance, from 20 kHz to 6 MHz. 
   The set of pulses generated sequentially are transmitted from the office  12  on the line  10  and they are reflected by the defects  16 ,  18 , as well as by the line end  21 , i.e. by subscriber  14 . In fact, it is known that a telephone set which is not in operation is a pure reflector. 
   The reflections or echoes are detected and processed at the central office  12  according to the equipment and method represented on  FIGS. 2-8  and these processed echoes provide information about the line  10  and its defects. 
   In the simplified example represented on  FIG. 2 , the vertical line  22  on the left represents the frequency and the diagram is, in the horizontal direction, separated in five zones  24 ,  26 ,  28 ,  30 , and  32  from the left to the right. The zone  24  corresponds to the transmission of pulses at the central office  12 . The zone  26  represents the path of the pulses from the central office  12  to the subscriber  14  and back from the subscriber  14  to the central office  12 . The zone  28  represents the processing of the received echoes, each processing corresponding to a given band-width. The zone  30  represents the sum ot the echoes which are processed in zone  28  and the zone  32  represents the spectrum of the transmitted, filtered and reconstructed pulses. 
   In the simplified example, three pulses  40 ,  42 ,  44  are provided. The pulse  40  has a spectrum which covers the low frequencies, the pulse  42  has a spectrum which covers the medium frequencies and the pulse  44  has a spectrum which covers the high frequencies. For instance, the pulse  40  covers 12.5 kHz to 960 kHz, the pulse  42  covers 512 KHz to 2.6 MHz and the pulse 44 covers 1.65 MHz to 6.3 MHz. The frequency bands of these pulses are overlapping. More precisely, the higher part of the frequency band of pulse  40  overlaps with the lower part of the frequency band of pulse  42  and the higher part of the frequency band of the pulse  42  overlaps with the lower part of the frequency band of pulse  44 . 
   Moreover, the overlapping portions of the spectra of the pulses are such that, after reflection, reception and processing, the amplitude of the added over-lapped parts equals the amplitude of the non-overlapping parts. In other words, after reception, the spectrum of all the pulses is practically flat, as shown on zone  32  of FIG.  2 . 
   For instance, the higher part  40   1  of the frequency band of pulse  40  varies as a sine function and the overlapping part of the lowest portion  42   1  of the frequency band of pulse  42 , varies as a cosine function. As in zone, or step,  30 , as explained herein after, the squares of the amplitudes of the spectra are added, the overlapping regions  40   1 ,  42   1  provide after the final processing: sin 2 +cos 2 =1. 
   The method provides also, on the transmission side, a gain (amplification or attenuation) for each pulse. For the sake of simplicity of the drawing, only the amplification and attenuation for pulse  42  have been represented. To the pulse  42  corresponds a gain represented by an amplifier  48 , and on the receiving side (zone  28 ), the inverse gain  50  is provided in order that the resulting spectra (zone  32 ) be flat as explained herein above. 
   Each gain may be different from one bandwidth to another bandwidth, in order to comply with the requirements of the standard which may impose different constraints on the admissible maximum power for different bandwidths. 
   On the receiving side, for each bandwidth, a processing is performed to drastically reduce the noise: a synchronous averaging  52 , followed by a matched filtering  54  and a denoising  56 . Each signal at the corresponding bandwidth, after having been processed by the synchronous averaging  52 , the matched filtering  54  and the denoising  56 , is submitted to a reconstruction step (zone  30 )  60  which sums the outputs of the processing steps. Because of the matched filter properties, the summation of the outputs of the processing steps is equivalent in the frequency domain, to the following equation: 
             ∑   i             ⁢           ⁢              P   i     ⁢           ⁢     (   f   )            2       =   1     ,       
 
where P i (f) is the transfer function for each pulse.
 
     FIG. 3   a  shows the variation with time (in abscissa) of the pulse  40 . The curve  62  corresponds to the real part of the pulse, the curve  64  represents the variation of the imaginary part, and the curve  66  represents the envelope of the pulse. 
   The diagram of  FIG. 3   b  shows the spectrum of the pulses  62  and  64 , i.e. the Fourier transform of said pulses. As shown, the spectrum, which extends from 20 kHz to 200 kHz presents a flat part  68 , a raising edge  70  and a falling edge  72 . As mentioned before, the falling edge has, for instance, the shape of a sine function. 
   The diagrams of  FIG. 4  represent the signals obtained after synchronous averaging  52  and before matched filtering  54 . 
     FIG. 4   a  represents the variation with time of the real part of the signal for the pulses  44  at high frequencies, and  FIG. 4   b  represents the variation with time of the imaginary part for the signal at the output of a synchronous averaging corresponding to the same high frequency pulses  44 . 
   The diagrams of  FIG. 4   c  and  FIG. 4   d  correspond, respectively, to the real and imaginary parts of the complex signal obtained after synchronous averaging for the medium frequency pulses and  FIG. 4   e  and  FIG. 4   f  are diagrams corresponding also to the real and imaginary parts of the signal obtained at the output of the synchronous averaging  52  for the low frequency pulses. 
     FIG. 4   a  and  FIG. 4   b  show that, for high frequencies, the echoes present a pulse  80 . This pulse corresponds to a defect  16  dose to the central office  12 , because the attenuation on the line increases sharply with the frequency and the distance. 
   For medium frequencies ( FIG. 4   c  and  FIG. 4   d ), the diagram shows two echoes  82  and  84  corresponding to defects  16  and  18  and, for low frequencies ( FIG. 4   e  and  FIG. 4   f ), the diagram shows several echoes corresponding to defects  16 ,  18 , and to the line end  21 . It is recalled that, in this example, the load coil defect  20  is not present. 
   The diagrams of  FIG. 5  are similar to the diagrams of FIG.  4 . but they represent the echoes obtained after the matched filtering  54 . The matched filtering comprises a step of correlating the received pulse with the transmitted pulse. This matched filtering provides a further sharp decrease of the noise, as shown by comparison of the diagrams of  FIG. 4   a  to  FIG. 4   f  with the corresponding diagrams of  FIGS. 5   a  to  5   f.    
   A further reduction of noise, more particularly for the high frequency pulses is obtained with the denoising  56 . This further noise suppression comprises a step of determining a threshold below which the values of signal and noise are set to zero, only the echoes which are above this threshold being taken into account. 
   The threshold is, in an example, determined by an estimation of the noise at the end (on the right of diagrams of  FIG. 5 ) of the signals obtained after matched filtering. In fact, the ending time corresponds to the end of the line, at the subscriber&#39;s location, for which no signal can be detected in medium and high frequencies; therefore, the signal end corresponds, in practice, exclusively to noise for high and medium frequencies. The noise is estimated by the variance of the signal at said signal end and the threshold is determined by multiplying the square root of this variance by a predetermined factor, for instance 2. More precisely, the noise variance is estimated after the line end echo, i.e. on a noise alone segment. 
   It is to be noted that the denoising is limited to the signals which appear after the last echo. No denoising is performed on signals appearing before the last echo. 
   The result of the denoising appears on the diagrams of  FIG. 6   a  to  FIG. 6   f  which are similar to the diagrams of  FIG. 4   a  to  FIG. 4   f.    
     FIG. 7  is a diagram showing the variation with time of the signal after the summing of the outputs of the processing step for the three frequency bands (output of adder  60 ). 
   This diagram shows that the time domain reflectometry method of the invention provides, in this example, three echoes  96 ,  98  and  100 . The first echo corresponds mainly to defect  16 , the second echo to defect  18  and the third to the line end  21 . 
   On the diagram of  FIG. 8 , the abscissa is the time t and the ordinate is the frequency f. It can be seen that, to the three lines  96 ,  98 ,  100 , correspond lines  102 ,  104  and  106 . The line  102  extends on the whole frequency band, the line  104  corresponds to the medium frequencies and the last line  106  is limited to low frequencies. 
   Therefore, the diagrams obtained with FIG.  7  and  FIG. 8  show that the method according to the invention provides, with only one measurement, information about the properties of the line and the defects, more particularly about the location of such defects. 
   More generally, the method according to the invention provides directly an estimation of the reflectometric impulse response of a line, with a good time accuracy and a high signal-to-noise ratio at any distance.