Abstract:
A low voltage differential I/O device and method is modeled using voltage sources and voltage dividers, rather than the current source and sink model of the prior art. In an exemplary implementation, a driver includes two pairs of transistors coupled between voltage sources, each pair associated with a respective logic state. Depending on the logic state to be signaled, one pair of transistors is driven strongly while the other pair is turned off. A differential voltage is established across the true and complement signal lines, the polarity of which is determined by which pair of transistors is driven, and the magnitude of which is readily determined by voltage division of the voltage sources across known resistances. The driver of the invention offers stable and low impedance across both logic states and common mode. Moreover, active devices and feedback are not required to establish a common mode voltage or impedance as in the prior art.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to data transmission devices, and more particularly, to low voltage differential signaling devices having desirable impedance and performance characteristics. 
     2. Description of the Related Art 
     Low Voltage Differential Signaling (LVDS) is a new technology aimed at high performance data transmission applications. LVDS technology features a low voltage differential signal of about 330 mV centered about a common-mode voltage of about 1.2V, compared to differential signals of about 0.6V-1.0V for PECL and about 2V-5V for RS-422 interfaces. With such low voltage swings, LVDS devices offer theoretical data rates in excess of 400 Mbps and relatively low power consumption (generally about 1.2 mW). In addition, LVDS allows for both DC and AC differences in ground reference voltage between the generator and receiver. This is accomplished by requiring the receiver to have a large input common mode range. Two key industry standards define LVDS: ANSI/TIA/EIA-644 and IEEE 1596.3. 
     Conventionally, LVDS drivers have been modeled as current mode devices, as illustrated in FIG.  1 . As shown, the desired differential voltage V diff  is achieved by causing a controlled current I to pass through termination resistor R term  (typically about 100 Ω) at the receiver (via transmission lines having impedance Z T ). Accordingly, the desired differential voltage V diff  between the true and complement signal lines  102  and  104  is determined as the product of R term  *I. Thus, given a known termination resistance R term , it is a matter of designing a circuit that will cause the correct current to flow through it and yield the desired voltage V diff . In most conventional LVDS drivers, this is done by adjusting the conductive bias on active current source devices such as FETs. Signaling is achieved by changing the polarity of the current I flowing through the termination resistor, and thus the polarity of V diff . Accordingly, much care has to be taken to properly design and implement biasing schemes. 
     In addition to required voltages to signal logic states, LVDS includes requirements concerning the “null” logic state, or “common-mode.” For example, LVDS specifies that driver impedances Z 1  and Z 2  should be about the same, relatively constant and matched to the load, independent of the logic state, so that common-mode noise that propagates backward and reflects off the drivers does not get converted into differential noise by different reflection coefficients on the true and complement signal lines. Moreover, a “common-mode” reference needs to be established and stably maintained. 
     Modeling LVDS drivers as current sources and sinks has many drawbacks and leads to many challenges. 
     For example, a current source and sink model, by itself, does nothing to set the common mode reference. Accordingly, as illustrated in FIG. 2, a reference voltage V cm  must be additionally provided with moderate impedance internal connections Rc 1  and Rc 2  thereto. The addition of Rc 1  and Rc 2  sets the common mode voltage and lowers the output impedance at the expense of increased power and complexity. If Rc 1  and Rc 2  are to provide perfect back termination (e.g. 100 ohms), then the current sources must provide twice the output current (i.e. doubling the power). Further, the demands on the voltage reference V cm  are not trivial, and cannot be integrated with the buffer without a prohibitive increase in power or area (for decoupling). Also, common mode testing requires that the V cm  supply both source and sink current. Therefore, V cm  cannot be generated without some degree of shunt regulation, thus wasting additional power. Lastly, this approach could not be used for bidirectional signaling since the V cm  supply would not allow input common mode to vary. 
     As another example, the common mode driver impedance according to the current source model becomes high and different than the driver impedance during signaling and as a result the common mode output voltage. So, as illustrated in FIG. 3, the prior art drivers typically add active feedback circuitry  302  to adjust the common mode voltage. However, this effectively lowers only the DC common mode driver impedance, at high frequency the delay in the circuit will result in reflections and additional artifacts. 
     Also, although adding such active devices can improve driver DC impedance, such a driver configuration is limited to point-to-point signaling. In other words, the transmission line associated with the driver can&#39;t be adapted to do bidirectional signaling with this configuration. This is because the active feedback could not be used to set the input impedance. 
     Another problem with the conventional driver of the prior art approach is that the driver impedance is high and not matched to either the transmission lines or the parasitic inductors, such as L 1  and L 2  in FIG.  3 . The high impedance output will cause reflections of single ended or differential noise when driving transmission lines longer than ¼ wavelength of the highest frequency noise source. These reflections will then interfere with unrelated data at the receiver. When driving short transmission lines, the high output impedance will allow L 1  and L 2  to resonate with the output capacitance. 
     TMDS attempts to solve some of the above-described problems with conventional LVDS drivers. The TMDS approach is to try to control the input impedance of the driver. However, like FIG. 3, the TMDS driver has a high output impedance and thus has similar problems with respect to reflecting noise and damping parasitic impedance. 
     Accordingly, there remains a need in the art for a low-voltage differential I/O device and method the does not need voltage references to set common mode voltages and provides constant impedance in all logic states without the need for active feedback circuitry. The present invention fulfills this need, among others. 
     SUMMARY OF THE INVENTION 
     Accordingly, an object of the present invention is to effectively overcome the above-described problems of the prior art, among others. 
     Another object of the present invention is to provide a low voltage differential I/O device and method that does not rely on voltage references. 
     Another object of the present invention is to provide a low voltage differential I/O device and method that does not require active devices and feedback to set common mode voltages. 
     Another object of the present invention is to provide a low voltage differential I/O device and method that provides constant impedance for each logic state. 
     Another object of the present invention is to provide a low voltage differential I/O device and method that achieves higher signaling rates over conventional LVDS devices. 
     Another object of the present invention is to provide a low voltage differential I/O device and method that can detune output inductance. 
     To achieve these objects and others, a low voltage differential I/O device and method according to the invention is modeled using voltage sources and voltage dividers, rather than the current source and sink model of the prior art. In an exemplary implementation, a driver includes two pairs of transistors coupled between voltage sources, each pair associated with a respective logic state. Depending on the logic state to be signaled, one pair of transistors is driven strongly while the other pair is turned off. A differential voltage is established across the true and complement signal lines, the polarity of which is determined by the pair of transistors that is driven, and the magnitude of which is readily determined by voltage division of the voltage sources across known resistances. The driver of the invention offers stable and low impedance across both logic states and common mode. Moreover, active devices and feedback are not required to establish a common mode voltage or impedance as in the prior art. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     These and other objects and advantages of the present invention, along with the best mode for practicing it, will become apparent to those skilled in the art after considering the following detailed specification, together with the accompanying drawings wherein: 
     FIG. 1 illustrates the current source and sink model of the low voltage differential driver of the prior art; 
     FIG. 2 illustrates a technique for setting common mode voltage in the current source and sink model of the low differential driver of the prior art; 
     FIG. 3 illustrates a technique for setting common mode driver impedance in the current source and sink model of the low differential driver of the prior art; 
     FIG. 4 illustrates a model of the low differential driver of the present invention; 
     FIG. 5 illustrates an equivalent circuit of the low differential driver of the present invention; 
     FIG. 6 illustrates an exemplary implementation of the low differential driver of the present invention; 
     FIG. 7 illustrates another exemplary implementation of the low differential driver of the present invention; and 
     FIGS. 8-A and  8 -B illustrate alternative models of the low differential driver of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     The principles of the invention will now be described with reference to FIG.  4 . As shown, a low-voltage differential I/O driver of the present invention is modeled using voltage sources rather than current sources and sinks. Specifically, voltage sources V 1  and V 2  are used to establish a voltage differential voltage V diff  between the true and complement signal lines  102  and  104  across R term . 
     The above model yields an equivalent circuit as shown in FIG.  5 . As should be apparent, if the back termination is ideally matched to R term , the desired differential voltage V diff  is achieved in accordance with a standard voltage division of the difference between V 1  and V 2 . Specifically, the desired differential voltage V diff  between the true and complement signal lines  102  and  104  is determined as (V 1 −V 2 )/2. 
     FIG. 6 illustrates an example of a circuit that can be used to implement the model LVDS driver in accordance with the invention. 
     In this example, driver  602  includes PFETs Q 1  and Q 3  having a width/length (W/L) of about 600, NFETs Q 2  and Q 4  having a W/L of about 300, resistors Rd 1 , Rd 2  and Rd 3  having a resistance of about 150 ohms, with V dd  being about 2.5 V and V sso  being about 0V. Moreover, resistor R term  has a resistance between 100 to 120 ohms, and Zt 1  and Zt 2  are transmission line impedances, such as PCB traces, of about 60 ohms each. 
     The parasitic capacitors and inductors are shown in FIG. 6 to provide a practical understanding of the invention to those skilled in the art. However, such parasitic devices can be effectively ignored for most purposes in the normal operating ranges of the present invention, where capacitors can be seen as open circuits and inductors can be seen as short circuits, except for certain aspects that will be explained in more detail below. 
     In the exemplary driver circuit shown in FIG. 6, when a differential signal having a first logic state, for example a “positive state”, is desired to be transmitted, input D+ is set to a positive voltage (e.g. Vdd or 2.5V) and input D− is set to a zero voltage (e.g. Vss or 0V). This causes transistors Q 1  and Q 4  to turn on and transistors Q 2  and Q 3  to turn off, thus causing resistor Rd 2  to be oriented in a conduction path between nodes A and B through transistors Q 1  and Q 4  such that a positive differential voltage of about 330 mV from common mode, is established between true and complement signal lines  102  and  104  due to the positive voltage drop across resistor Rd 2 . 
     It should be noted that the transistors in the driver of the present invention are designed so as to be driven strongly into conductance (i.e. “triode” or “linear” mode of operation), and not just saturation. This contrasts with the prior art, where circuits aim at providing a carefully controlled saturation-mode current through a termination resistor close to the receiver. In the present invention, therefore, transistors Q 1 , Q 2 , Q 3  and Q 4  are effectively operated as switches rather than voltage-controlled current sources as in the prior art, and the desired differential voltage is effectively established as a voltage division between resistors Rd 2  and R term . 
     When a differential signal having a second logic state, for example a “negative state”, is desired to be transmitted, input D+ is set to a zero voltage (e.g. Vss or 0V) and input D− is set to a positive voltage (e.g. Vdd or 2.5V). This causes transistors Q 2  and Q 3  to turn on and transistors Q 1  and Q 4  to turn off, thus causing resistor Rd 2  to be oriented in a conduction path between nodes A and B through transistors Q 2  and Q 3  such that a negative differential voltage of about 330 mV from common mode is established between the true and complement signal lines  102  and  104  due to the negative voltage drop across resistor Rd 2 . 
     It should be noted that the driver impedances Z 1  and Z 2  of driver  602  are going to be the same for all logicstates as well as for common mode due to the operation of transistors Q 1 , Q 2 , Q 3  and Q 4  and the constant impedances provided by Rd 1 , Rd 2  and Rd 3 . Moreover, the desired common mode voltage is easily established in both logic states by operation of the driver itself when either transistors Q 1  and Q 4  are turned on and Q 2  and Q 3  are turned off or transistors Q 2  and Q 3  are turned on and Q 1  and Q 4  are turned off. Accordingly, it should be apparent that the common mode voltage Vcm will be determined by the difference between the voltage at nodes A and B regardless of which pairs of transistors Q 1 /Q 4  and Q 2 /Q 3  are turned on and which are turned off. 
     Another advantage of the driver  602  of the present invention is that driver output impedance problems discussed in connection with FIG. 3 above, are substantially reduced. In particular, for long transmission lines, because the output impedance is ideally matched with the transmission lines, noise reflections are reduced. For short transmission lines, energy arising from stimulation of the parasitic inductors Lp 1  and Lp 2 , is absorbed by the back termination Rd 2  and the parasitic capacitance, thus providing sufficient dampening against ringing on the transmission lines. 
     Yet another advantage of the present invention is that the resistor Rd 2  of this configuration can be used as a termination resistor when the transmission lines are being used for bidirectional signaling. For bidirectional signaling mode, inverters D 1 , D 2 , D 3 , and D 4  are driven so as to turn transistors Q 1 , Q 2 , Q 3  and Q 4  off, which leaves a parallel resistance of R term and Rd 2  between the true and complement signal lines  102  and  104 , and allows the voltage between the lines to float to around to the common mode voltage as established by the generator. Accordingly, Rd 2  in this mode acts as a receiver termination resistor R term . 
     FIG. 7 illustrates another example of an LVDS driver  702  in accordance with the invention. In this example, capacitors Cd 1  and Cd 2  are further provided to provide extra charge to turn the devices on, thus yielding improved switching speeds. 
     As shown, capacitor Cd 1  is connected between the gate of transistor Q 1  and the drain of transistor Q 3 , and capacitor Cd 2  is connected between the gate of transistor Q 3  and the drain of transistor Q 1 . Capacitors Cd 1  and Cd 2  each have a capacitance of about 1 pF and cause the parasitic capacitances to charge up at lower currents, thus reducing the delay in switching speeds caused by this factor. 
     FIGS. 8-A and  8 -B illustrate alternative models for an LVDS driver in accordance with the invention. In this alternative, additional segments are provided to more easily match the impedance of the driver to the load, for example. 
     Specifically, as shown in FIG. 8-A, alternate resistances R 2   a,  R 2   b,  and R 2   c  are provided in parallel with R 2 . The selection of the overall resistance that is used to establish V diff  is thus made by controlling the selection signals SEL-a, SEL-b and SEL-c that are connected to the gates of NFETs Qa, Qb and Qc. For example, if all signals SEL-a, SEL-b and SEL-c are set so that NFETs Qa, Qb and Qc turn off, the resistance is simply R 2 . As another example, if signals SEL-b and SEL-c are set so that NFETs Qb and Qc turn off, while signal SEL-a is set so that NFET Qa turns on, the resistance for establishing V diff  is the parallel resistance of R 2  and R 2   a.    
     As shown in FIG. 8-B, alternate resistances are provided to variably adjust resistances R 1  and R 3 . In this example, R 1   a  is preferably substantially equal to R 3   a,  R 1   b  is preferably substantially equal to R 3   b  and R 1   c  is preferably substantially equal to R 3   c.  Moreover, selection signal SEL-b is commonly connected to the gates of PFET Q 1   b  (through an inverter) and NFET Q 3   b,  and selection signal SEL-c is commonly connected to the gates of PFET Q 1   c  (through an inverter) and NFET Q 3   c.  As should be apparent, therefore, the selection of the overall resistance that is used to establish resistances R 1  and R 2  is thus simultaneously made by controlling the selection signals SEL-b and SEL-c. For example, if signals SEL-b and SEL-c are set so that PFETs Q 1   b  and Q 1   c  and NFETs Q 3   b  and Q 3   c  turn off, the resistances of R 1  and R 3  are the sum of R 1   a,  R 1   b,  R 1   c  and R 3   a,  R 3   b  and R 3   c,  respectively. As another example, if signal SEL-c is set so that PFET Q 1   c  and NFET Q 3   c  turn on, while signal SEL-b is set so that PFET Q 1   b  and NFET Q 3   b  turn off, the resistances R 1  and R 3  are the sum of R 1   a,  R 1   b  and R 3   a,  R 3   b  respectively. While this tuning of the resistors is shown using the addition of series resitors, it could also been done using a combination or parallel resistors. 
     Although the present invention has been described in detail with reference to the preferred embodiments thereof, those skilled in the art will appreciate that various substitutions and modifications can be made to the examples described herein while remaining within the spirit and scope of the invention as defined in the appended claims.