Abstract:
A noise reduction filter is inserted between the source and non-linear transmission line (NLTL) in a frequency multiplier to improve phase noise performance. The noise reduction filter is suitably coupled directly to the input of the NLTL. The noise reduction filter and the output BPF are suitably low complexity filters.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     This invention relates to Non-Linear Transmission Line (NLTL) Frequency Multipliers, and more particularly to noise reduction for NLTL Frequency Multipliers. 
     2. Description of the Related Art 
     Microwave systems often require high frequency input signals. Frequency multipliers are used to translate a low frequency input signal to a desired higher frequency. Frequency multipliers include a source of the low frequency input signal, a comb generator that produces output signals at multiple harmonics of the input signal, a band pass filter (BPF) that selects one of the harmonics and amplifier. 
     Conventional comb generators produce the harmonics using step recovery diodes (SRD). SRD implementations generally accept input signals over a narrow range of frequencies and power levels, thereby limiting user selection of harmonics spacing and frequency range. In addition, SRD implementations can introduce substantial phase noise. 
     A new family of comb generators based on nonlinear transmission line (NLTL) technology has demonstrated improved phase noise and a wider input power range. NLTL generators create output harmonics through the nonlinear nature of propagation within the device, avoiding exposure to recombination and shot noise that is prevalent within step recovery diodes. The aggregate effect of this new technology is that the residual phase noise is dramatically better—NLTL comb generators are exhibiting at least a 20 dB improvement over their SRD counterparts. 
     Referring now to  FIG. 1 , a nonlinear transmission line (NLTL)  10  is a transmission line formed from a periodic structure of series inductors  12  and variable shunt capacitors  14 . The variable shunt capacitors are suitably voltage sensitive Schottky varactor diodes. The capacitance of a reverse biased Schottky diode is voltage dependent such that the capacitance at low reverse bias is much greater than the capacitance at high reverse bias. An input signal  16  propagating on the equivalent transmission line made with varactors experiences a propagation velocity that is voltage dependent. A signal that transitions from low to high voltage will be compressed in time as the initial low voltage portion of the signal travels down the line slower than the later, higher voltage portion of the signal. Consequently, the higher voltage portion of the waveform “catches up” with the lower voltage portion of the step, resulting in increasing the edge speed of the low to-high transition. This sharper rising edge waveform produces an output signal  18  that is rich in signal harmonics in the frequency spectrum. A more complete description of a NLTL is provided in Mark J. Rodwell et al. “GaAs Nonlinear Transmission Lines for Picosecond Pulse Generation and Millimeter-Wave Sampling” IEEE Transactions on Microwave Theory and Techniques, Vol. 39, No. 7, July 1991, pp. 1194-1204 and Wenjia Tang et al. “Low Spurious, Broadband Frequency Translator using Left-Handed Nonlinear Transmission Line” IEEE Microwave and Wireless Components letters, Vol. 19. No. 4, April 2009, pp. 221-223, which are hereby incorporated by reference. 
     Referring now to  FIGS. 2 and 3 , a frequency multiplier  20  includes a source  22  that supplies an input signal  24  (frequency-domain representation  26 ) at a frequency F o , a NLTL  27  that propagates the input signal nonlinearly to produce a sharp rising edge waveform  28  with multiple harmonics  30  of the input signal in the frequency domain, a band pass filter (BPF)  32  that selects one of the harmonics  33  (NF 0 ) as an output signal  34  and an amplifier  35  that amplifies the output signal  34 . A more complete description of a frequency multiplier using NLTL technology is provided in U.S. Pat. Nos. 7,462,956 and 7,612,629, which are hereby incorporated by reference. 
     Source  22  typically includes an oscillator  36  that generates input signal  24  at a given frequency F 0 . The amplitude level of input signal  24  must match the input range of the NLTL. Typical sources generate the input signal  24  at a fixed amplitude that does not match the NLTL. Typically, the input signal needs to be amplified. In an embodiment, source  22  includes an amplifier  37  that provides a fixed amount of gain, an input attenuator  38  that attenuates input signal  24  so that its amplitude lies in the linear region of amplifier  37  and an output attenuator  40  that attenuates the amplified signal to provide level adjustment to match the input range of the NLTL. Other source configurations are possible. 
     BPF  32  has a pass band  42  that is approximately centered at the desired harmonic NF 0  and sufficiently wide to pass harmonic  33  and side bands  44  that provide sufficient attenuation to reject all other harmonics. Typically, the side bands  44  must satisfy a specified side band rejection requirement  46  (e.g. −40 dB attenuation) at the adjacent harmonics. Filter “Q” determines the width of the pass band  42  and how sharp side bands  44  transition from the pass band level to a high attenuation level. A high Q filter transitions quickly and a low Q filter rolls off slowly. A high Q filter can provide greater side band rejection but is more complex (i.e. a higher order filter), hence costly. Generally speaking, a circuit designer would prefer to select the lowest Q filter that satisfies the side band rejection requirement. 
     SUMMARY OF THE INVENTION 
     The following is a summary of the invention in order to provide a basic understanding of some aspects of the invention. This summary is not intended to identify key or critical elements of the invention or to delineate the scope of the invention. Its sole purpose is to present some concepts of the invention in a simplified form as a prelude to the more detailed description and the defining claims that are presented later. 
     The present invention provides a frequency multiplier based on NLTL technology with reduced phase noise. 
     In an embodiment, a frequency multiplier comprises a source configured to generate an input electrical signal at an input frequency and amplitude. A noise reduction filter is configured with a pass band to pass the electrical signal at the input frequency and a rejection band to reject a band of low frequencies below the input frequency. A non-linear transmission line (NLTL) is configured to time delay the electrical signal as a function of amplitude to generate electrical signals at integer multiples of the input frequency F 0 . A band pass filter (BPF) is configured with a pass band to pass one of electrical signals at a particular integer multiple N of the input frequency F 0  as a frequency-multiplied electrical signal and side bands to reject all other multiples of the input frequency. An amplifier is configured to amplify the frequency-multiplied electrical signal. Incorporation of the noise reduction filter reduces phase noise in both the main lobe of the frequency-multiplied electrical signal and a side lobe. 
     In an embodiment, the NLTL is directly coupled to the output of the noise reduction filter. There is no intervening electrical component between the noise reduction filter and the NLTL. 
     In an embodiment, the noise reduction filter is either a HPF or a BPF. 
     In an embodiment, the noise reduction filter is at most a 5 th  order filter and the BPF is at most a 7 th  order filter. 
     In an embodiment, a cut-off frequency Fc that separates the pass band and the rejection band of the noise reduction filter lies below the input frequency offset by half the bandwidth of the input electrical signal. The rejection band provides at least 10 dB of rejection at a frequency Ft where 1/f noise equals the wide band noise floor. The width of the BPF&#39;s pass band is between 2 to 5 percent of the selected harmonic NF 0 . 
     These and other features and advantages of the invention will be apparent to those skilled in the art from the following detailed description of preferred embodiments, taken together with the accompanying drawings, in which: 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1 , as described above, is an equivalent circuit of a NLTL; 
         FIG. 2 , as described above, is a schematic of a NLTL frequency multiplier; 
         FIG. 3 , as described above, is a diagram illustrating the manipulation of an input signal by the NLTL to create multiple harmonics and the selection of the frequency multiplied signal by the BPF; 
         FIG. 4  is a plot of the single sideband phase noise of the frequency multiplied output signal; 
         FIGS. 5   a  and  5   b  are plots of input and output noise spectra for the frequency multiplier; 
         FIG. 6  is an embodiment of a noise-reduced NLTL frequency multiplier; 
         FIG. 7  is a schematic of a High Pass Filter (HPF); 
         FIG. 8  is a drawing of the HPF response overlaid on the input noise spectrum; 
         FIG. 9  is a plot of the noise spectrum of the noise-reduced frequency multiplied output signal; and 
         FIG. 10  is a plot of the noise spectrum of a noise-reduced frequency multiplied output signal in which the HPF and NLTL are separated by an attenuator. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     The present invention describes a frequency multiplier based on NLTL technology with reduced phase noise. 
     Single-sideband phase noise is the phase instability of the local oscillator (e.g. frequency multiplier) measured in the frequency domain. It is the most commonly used measurement of phase noise. The single side band noise is defined as the ratio of the carrier power to the noise power in a 1 Hz bandwidth, expressed in dBc/Hz, at the given frequency offset, Δf, from the carrier (e.g. the input signal at F 0  or the selected harmonic NF 0 ). In general, phase noise falls off exponentially with frequency offset. Thermal noise caused by random collisions of charge carriers with atoms of the lattice found in metal conductors and resistive materials is the primary source of phase noise in microwave circuits. The oscillator, attenuators and amplifier all contribute to thermal noise. A spectrum analyzer can be used to measure phase noise if the oscillator has no amplitude noise modulation and the phase noise of the spectrum analyzer&#39;s oscillators is less than the measured oscillator. Measurement of phase noise near the carrier frequency (&lt;100 Hz offset) is time consuming and very difficult, hence typically not done. Similarly, measurement of phase noise at large offsets (&gt;10 MHz offset) from the carrier requires expensive spectrum analyzers, and is typically not done. 
     Single-sideband phase noise is important to a local oscillator (e.g. a frequency multiplied source) for a microwave exciter/receiver. Phase noise affects receiver selectivity in a multi-signal environment. Multiplication (or mixing) in the time domain is equivalent to convolution in the frequency domain. Therefore phase noise of one signal will be superimposed upon the other signal during the frequency conversion. Of most interest is the local oscillator phase noise mixing onto the incoming signal of interest. Consequently, phase noise considerations play a major role in the selection of components and circuit architecture. 
       FIG. 4  is a plot of the single sideband phase noise  100  for a conventional frequency multiplier based on NLTL technology as illustrated in  FIGS. 1-3  above in which the harmonics are separated by about 500 MHz. Our interest is directed to local oscillators such as provided by the frequency multiplier with very stringent phase noise requirements for microwave exciters/receivers. We measured the phase noise of the frequency multiplier circuit from approximately 100 Hz out to an offset of approximately 100 MHz. This test revealed a “side lobe”  102  between approximately 40 MHz and 100 MHz in which the phase noise unexpectedly and markedly increased. Although not directly apparent from this plot of phase noise, the main lobe  104  of the selected harmonic included an additional phase noise component as well, approximately 10 dB at 100 Hz. This was only discovered after we inserted a noise reduction filter before the NLTL to reduce side lobe  102 . 
     Although the source of the phase noise that creates the side lobe  102  and adds to the phase noise of the main lobe  104  is unknown, we hypothesize that it is caused by a nonlinear interaction of the NLTL with 1/f noise that is present in all electrical components. The exact origins of 1/f noise are unknown. However, 1/f noise is a time dependent noise term that is a function of how long something is observed. The closer the observation is to the carrier the longer the observation time, hence the larger the phase fluctuations. 
     Referring now to the figures,  FIG. 5   a  depicts a hypothesized input noise spectrum  110  whose interaction with the NLTL may produce the observed output noise spectrum  112  shown in  FIG. 5   b  as double-sided phase noise about the carrier frequency NF o . As hypothesized, input noise spectrum  110  includes the phase noise  113  of the source (e.g. oscillator, attenuators and amplifier) about the input frequency, a wideband noise floor  114  that is set by thermal noise as well and a 1/f noise component  116 . 1/f noise is a low frequency noise that would not be expected to impact the phase noise of the output carrier frequency (e.g. selected harmonic NF 0 ) at the output of the NLTL. However, our hypothesis is that somehow the 1/f noise is interacting with the NLTL to superimpose a significant measure of phase noise on the main lobe  118  and create a side lobe  120  in the observed output noise spectrum  112 . 
     The frequency response  122  of a standard low-Q BPF (no greater than 7 th  order) used in frequency multipliers is overlaid on the output noise spectrum  112 . The width of the pass band is such that the side lobes  120  are not attenuated. A typical width of the pass band being 2-5% of the output carrier signal frequency. As a result, the phase noise in the side lobes contributes to the overall phase noise of the output carrier signal and local oscillator. Furthermore, the BPF cannot remove the phase noise component that has been added to the main lobe at the carrier signal. One approach to reducing the phase noise would be to use a high-Q BPF having a frequency response  124 . The illustrated frequency response would require a 5 th  order or higher BPF with a pass band bandwidth of a fraction of 1%. The high-Q BPF would attenuate the side lobe contributions but could not remove the main lobe component. Furthermore, designers typically wish to avoid the additional complexity and cost of high-Q BPFs. 
     Referring now to  FIG. 6 , in accordance with an embodiment of the invention a noise reduction filter  130  is added to a frequency multiplier  132  between source  134  and NLTL  136 . Noise reduction filter  130  is configured to pass the input signal frequency and reject low frequency 1/f noise. Noise reduction filter  130  may be a high pass filter (HPF) or another BPF. Measurements of the single side phase noise have shown that the noise reduction filter  130  attenuates both the side lobe and main lobe phase noise components. The noise reduction filter  130  provides better performance than even the high-Q BPF at the output, and the total complexity and cost of a low-order noise reduction filter  130  and a low-order BPF  138  is less than a single high-order BPF at the output.  FIG. 7  is a schematic diagram of an embodiment of noise reduction filter  130  configured as a 5 th  order high pass filter (HPF). 
     Frequency multiplier  132  includes source  134  that supplies an input signal  140  at a frequency F 0 , noise reduction filter  130  that passes the input signal  140  while rejecting lower frequencies, NLTL  136  that propagates the input signal nonlinearly to produce a sharp rising edge waveform  142  with multiple harmonics of the input signal in the frequency domain, BPF  138  that selects one of the harmonics (NF 0 ) as an output carrier signal  143  and an amplifier  144  that amplifies the output carrier signal  143 . 
     Source  134  typically includes an oscillator  146  that generates input signal  140  at a given frequency F 0 . The amplitude level of input signal  140  must match the input range of the NLTL. Typical sources generate the input signal  140  at a fixed amplitude that does not match the NLTL. Typically, the input signal needs to be amplified. In an embodiment, source  134  includes an amplifier  148  that provides a fixed amount of gain, an input attenuator  150  that attenuates input signal  140  so that its amplitude lies in the linear region of amplifier  148  and an output attenuator  152  that attenuates the amplified signal to provide level adjustment to match the input range of the NLTL. Other source configurations are possible. 
     The output band pass filter  138  is designed to reject the adjacent comb sidebands to the required level for the given application with the widest possible pass band bandwidth (allowing for component and temperature variations) and keeping the filter order as low as possible for lowest size and cost. Typical filter bandwidths range in the 2%-5% with filter orders being at most 7 and typically in the 5 th  order range. 
     In order to maintain the amplitude of the selected harmonic, an absorptive band pass filter  138  is employed at the output of the comb generator to absorb rather than reflect input signal harmonics that can destructively cancel the required output signal. In a typical band pass filter, out of band signals are reflected back to the source due to a 0 dB return loss characteristic. These reflected signals can be out of phase and sub harmonically related with the output signal. The combined in phase and out of phase signals cancel each other and drastically lower the output signal amplitude. The same effect may be achieved with a reflective BPF  138  if the order of the Harmonic is a prime number, or possibly just an odd number. In this case, the combined in phase and out of phase signals will not cancel each other. 
     As shown in  FIG. 8 , the frequency response  150  of a HPF (the noise rejection filter  130 ) is overlaid on the input noise spectrum  110 . Frequency response  150  includes a pass band  152  and a rejection band  154  that are separated by a cutoff frequency Fc. A frequency Ft is defined as the frequency at which the 1/f noise  116  equals the wide bandnoise floor  114 . Rejection band  154  provides sufficient attenuation to satisfy a 1/f noise rejection requirement (e.g. at least 10 dB) at a frequency Ft where 1/f noise equals the wide band noise floor. 
     Cutoff frequency Fc must be chosen to pass the required frequency content  113  of the input signal  112  while rejecting the low frequency 1/f noise  116 . Regardless of the filter insertion loss, wanted signals must be above Fc while unwanted signals must fall below Fc. Inherent component and temperature variations will cause an uncertainty in the absolute value of Fc. If a cutoff frequency is chosen too close to the input signal frequency F 0 , filter component variations can cause the HPF stop band frequency to fall on the input frequency and attenuate the input signal. If the cutoff frequency Fc is chosen too close to the 1/f noise spectrum, than a higher order filter (more components) will need to be used to reject the low frequency 1/f noise. In most applications, there is a large range of HPF cutoff frequencies that can be chosen to pass the input signal while attenuating the low frequency 1/f noise while keeping the filter order low (minimum parts count) and accounting for known component variations. 
       FIG. 9  is a plot of the single sideband phase noise  160  for a noise reduced frequency multiplier based on NLTL technology of the type shown in  FIG. 6 . The insertion of a noise reduction filter in the frequency multiplier immediately prior to the NLTL eliminated the side lobe between 40 and 100 MHz and reduced the phase noise at 100 Hz by approximately 10 dB, this being a clear indicator of reduced phase noise in the main lobe. This was achieved with a 5 th  order HPF implementation of the noise reduction filter and a 5 th  order implementation of the absorptive BPF. 
       FIG. 10  is a plot of the single sideband phase noise  170  for a noise reduced frequency multiplier based on NLTL technology of the type shown in  FIG. 6 . However, in this example, an attenuator was placed between the noise reduction filter and the NLTL. The result was the reemergence of a “side lobe’  172  at about 20 MHz. This shows the importance of directly coupling the NLTL to the output of the noise reduction filter. 
     While several illustrative embodiments of the invention have been shown and described, numerous variations and alternate embodiments will occur to those skilled in the art. Such variations and alternate embodiments are contemplated, and can be made without departing from the spirit and scope of the invention as defined in the appended claims.