Abstract:
An improved Single-Stage Buck-Boost inverter (S 2 B 2  Inverter) is provided, using only three or four power semiconductor switches and two coupled inductors in a flyback arrangement. The inverter can handle a wide range of dc input voltages and produce a fixed ac output voltage. The inverter is well suited to distributed power generation systems such as photovoltaic and wind power and fuel cells, for standalone or grid connected applications. The inverter has a single charge loop, a positive discharge loop and a negative discharge loop.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS  
       [0001]     This application claims the benefit of U.S. provisional patent application 60/557,688, filed Mar. 31, 2004. 
     
    
     MICROFICHE APPENDIX  
       [0002]     Not Applicable.  
       TECHNICAL FIELD  
       [0003]     The present invention relates generally to power converters and more particularly to DC to AC buck-boost power inverters.  
       BACKGROUND OF THE INVENTION  
       [0004]     Increasing global energy consumption and noticeable environmental pollution are making renewable energy more important. Today, a small percentage of total global energy comes from renewable sources, mainly hydro and wind power. However, global energy consumption is expected to expand by 58% between 2001 and 2025. As more countries ratify the Kyoto Accord, an international agreement to reduce greenhouse gas (GHG) emissions, new power generation capacity can no longer be met by traditional methods such as burning coal, oil, natural gas, etc. Also, these traditional sources are predicted to last only about 100 to 200 years in the world. Nuclear power plants have experienced safety problems and disposal of nuclear waste remains a serious issue. These issues increase the importance of renewable energy.  
         [0005]     Energy from the wind, sun, water, waves, tides, etc., is renewable and essentially inexhaustible but the output from such sources is widely dispersed and generally sporadic, fluctuating dramatically with the weather and the seasons. Distributed generation (DG) technologies provide a potential solution of increasing electrical power generation capacity for renewable energy systems. Compared to large, centralized power grids, DG systems are usually small modular devices with increased security and reliability, and are generally close to electricity users, thus reducing the problems of power transmission and power quality issues due to very long transmission lines. DG systems often need dc-ac converters or inverters as an interface between their power sources and their typical single-phase loads. DG systems typically must deal with a wide range of input voltage variations due to the sporadic nature of the energy sources, which imposes stringent requirements on power inverters. Power inverters for small DG systems typically have the following requirements: (1) converting the variable incoming dc voltage into a fixed ac voltage with a fixed frequency; (2) ensuring output power quality with well controlled output frequency and low total harmonic distortion (THD); (3) providing electrical isolation and protection if necessary; and (4) low cost and high efficiency. DG systems are typically used to supplement the traditional electrical power grid and are often connected to the grid. In such cases, output power quality must meet specific standards, such as the interconnection requirements of IEEE 1547. For DG systems, the power grid source is strong enough to establish the output voltage waveform of inverters, thus the output current waveform and output power are often controlled objectives.  
         [0006]     Traditional single-phase full-bridge inverters  100 , as shown in  FIG. 1  do not have the flexibility of handling wide ranges of input voltage. They often require large, heavy line-frequency step-up transformers  102  when handling low voltage dc inputs.  
         [0007]     Examples of prior art two stage inverters are shown in  FIG. 2 ,  FIG. 3 ,  FIG. 4 , and  FIG. 5 .  
         [0008]     Interest in buck-boost inverters has grown notably with the development of sustainable DG energy systems in recent years, because buck-boost inverters can handle a wide range of input voltages, both lower and higher than the desired ac output voltage. Examples of prior art two stage buck-boost inverters are shown in  FIG. 6 ,  FIG. 7 , and  FIG. 8 .  
         [0009]     Compared to two-stage buck-boost inverters, most of single-stage buck-boost inverters present a compact design with a good performance-cost ratio, but they suffer from low power capacity and limited operation range imposed to dc sources. Several S 2 B 2  inverter topologies have been proposed in recent years. Examples of prior art single-stage buck-boost inverters are shown in  FIG. 9 ,  FIG. 10 ,  FIG. 11 ,  FIG. 12 ,  FIG. 13 ,  FIG. 14 , and  FIG. 15 . Some of them still have higher component count and more complicated operations, even compared with a two-stage inverter, and thus compromise their benefits. Others limit their applications by either requiring split dc sources ( FIG. 12 ) or imposing very high switching frequency ( FIG. 13 ,  FIG. 15 ) to demonstrate performance, or presenting low power ratings ( FIG. 15 ).  
         [0010]     Accordingly, an improved power converter having low power component count, wide input voltage range and improved performance, remains highly desirable.  
       SUMMARY OF THE INVENTION  
       [0011]     It is therefore an object of the present invention to provide an improved single-stage buck-boost inverter for handling a wide range of DC input voltage.  
         [0012]     The simple circuit topology of the present invention provides the possibility for a low cost and high efficiency power converter. The inverter has a low component count with only four power switches, four diodes, and a compact high frequency transformer. Compared to traditional buck inverters with line-frequency transformers, two-stage buck-boost inverters, and many of single-stage buck-boost inverters, both the cost and size are reduced, thereby presenting a more reliable and more economic design in small DG systems. Two current control schemes, DCM and CCM, are presented with amplitude modulation techniques.  
         [0013]     Accordingly, an aspect of the present invention provides a single-stage buck-boost inverter, comprising: an input for receiving DC power; an output operable to provide AC power; a first inductor; a first switching means to controllably connect said first inductor to said input; a first discharge loop for conveying power to said output; a second switching means to controllably connect said first inductor to said first discharge loop; a second inductor magnetically coupled to said first inductor; a second discharge loop for conveying power to said output; a third switching means to controllably connect said second inductor to said second discharge loop.  
         [0014]     In some embodiments, said first discharge loop is operable to convey power during a positive half cycle of output and said second discharge loop is operable to convey power during a negative half cycle of output.  
         [0015]     Some embodiments, further comprising a control means for controlling said first, second and third switching means so as to generate an AC power signal at said output.  
         [0016]     In some embodiments, said first switching means comprises a first semiconductor switch and a second semiconductor switch.  
         [0017]     In some embodiments, said second switching means comprises said first semiconductor switch and a third semiconductor switch and said third switching means comprises said second semiconductor switch and a fourth semiconductor switch.  
         [0018]     In some embodiments, each said first and second discharge loop further comprises an isolation diode to isolate said output from said input.  
         [0019]     In other embodiments said first switching means comprises one semiconductor switch.  
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0020]     Further features and advantages of the present invention will become apparent from the following detailed description, taken in combination with the appended drawings, in which:  
         [0021]      FIG. 1  is a schematic illustration of a prior art full bridge inverter;  
         [0022]      FIG. 2  is a schematic illustration of a prior art two stage inverter;  
         [0023]      FIG. 3  is a schematic illustration of a prior art two stage inverter;  
         [0024]      FIG. 4  is a schematic illustration of a prior art two stage inverter;  
         [0025]      FIG. 5  is a schematic illustration of a prior art two stage inverter;  
         [0026]      FIG. 6  is a schematic illustration of a prior art two stage buck-boost inverter;  
         [0027]      FIG. 7  is a schematic illustration of a prior art two stage buck-boost inverter;  
         [0028]      FIG. 8  is a schematic illustration of a prior art two stage isolated buck-boost inverter;  
         [0029]      FIG. 9  is a schematic illustration of a prior art single-stage buck-boost inverter;  
         [0030]      FIG. 10  is a schematic illustration of a prior art single-stage buck-boost inverter;  
         [0031]      FIG. 11  is a schematic illustration of a prior art single-stage buck-boost inverter;  
         [0032]      FIG. 12  is a schematic illustration of a prior art single-stage buck-boost inverter;  
         [0033]      FIG. 13  is a schematic illustration of a prior art single-stage buck-boost inverter;  
         [0034]      FIG. 14  is a schematic illustration of a prior art single-stage buck-boost inverter;  
         [0035]      FIG. 15  is a schematic illustration of a prior art single-stage buck-boost inverter;  
         [0036]      FIG. 16  is a block diagram of a typical wind energy system for use with the present invention;  
         [0037]      FIG. 17  is a schematic illustration of a first embodiment of a single-stage buck-boost inverter of the present invention;  
         [0038]      FIG. 18  is a schematic illustration of an approximate equivalent circuit of the charge mode of the embodiment of  FIG. 17 ;  
         [0039]      FIG. 19  is a schematic illustration of an approximate equivalent circuit of the positive half-cycle (PHC) discharge mode of the embodiment of  FIG. 17 ;  
         [0040]      FIG. 20  is a schematic illustration of an approximate equivalent circuit of the negative half-cycle (NHC) discharge mode of the embodiment of  FIG. 17 ;  
         [0041]      FIG. 21  is a graphical representation of exemplary inductor current and capacitor voltage;  
         [0042]      FIG. 22  is a graphical representation of exemplary unfiltered output current in a DCM scheme;  
         [0043]      FIG. 23  is a graphical representation of exemplary unfiltered output current in a CCM scheme;  
         [0044]      FIG. 24  is an illustration of an exemplary flyback transformer (magnetically coupled inductors) of the present invention;  
         [0045]      FIG. 25  is a block diagram of an exemplary control circuit of the present invention; and  
         [0046]      FIG. 26  is a schematic illustration of a second embodiment of a single-stage buck-boost inverter of the present invention. 
     
    
       [0047]     It will be noted that, throughout the appended drawings, like features are identified by like reference numerals.  
       DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT  
       [0048]     The present invention provides an improved Single-Stage Buck-Boost (S 2 B 2 ) inverter. Most S 2 B 2  inverters are derived from buck-boost dc-dc converter designs, where flyback principles are applied to transfer the energy from input side to output side. Two basic criteria are used to construct a buck-boost dc-dc converter, also called dc chopper. These criteria are: 1) an independent charge loop with respect to load, where an inductor or flyback transformer is usually used to store the energy; and 2) an independent discharge loop with respect to dc source, where the energy-storage element acts as the source to load.  
         [0049]     For a S 2 B 2  inverter, the first criterion guarantees no short circuit in dc link, thereby avoiding the dead-time problem as seen in traditional buck inverters. For grid-connected inverters, the second criterion decouples the ac output from the dc source when discharging, and therefore facilitates a sine wave output. For unidirectional grid-connected inverter systems, there are two additional criteria: 3) half-wave inversion; and 4) proper isolation of power flow from grid to dc source. In some cases for safety reasons, electric isolation is achieved by high-frequency transformers.  
         [0050]     Addressing the aforesaid four criteria, different topologies can be integrated by combining either: 1) one inductor, one charge loop, and two discharge loops; 2) one inductor, two charge loops, and one discharge loop, like the topology of  FIG. 15 ; or 3) two inductors, two charge loops, and two discharge loops, like the topologies of  FIGS. 9, 10 , and  12 .  
         [0051]      FIG. 17  is a schematic illustration of a first embodiment of the S 2 B 2  inverter  170  of the present invention. It consists of one charge loop  171  (T 1    180 , L 1    178 , and T 4    183 ) and two discharge loops  172 ,  173  (T 1    180 , L 1    178 , C  184 , D 3    176 , and T 3    182  for the positive half cycle  172 ; and T 4    183 , L 2    179 , D 2    177 , T 2    181 , and C  184  for the negative half cycle  173 ). The reverse power flow from grid  174  to source V s    175  is blocked by D 3    176  and D 2    177 . The energy-storage components, L 1    178  and L 2    179 , are the primary and secondary windings of a flyback transformer and have identical inductance (L) and number of turns. As is well understood in the art, the switches T 1    180 , T 2    181 , T 3    182  and T 4    183 , are controlled by a separate control circuit not illustrated in  FIG. 17 .  
         [0052]     Each of the functional loops  171 ,  172 ,  173  is associated with one of three switch operation modes. Charge loop  171  is used in charge mode, wherein switch T 1    180  and T 4    183  are on and switch T 2    181  and T 3    182  are off. An approximate equivalent circuit is shown in  FIG. 18  without the consideration of inductor copper loss and semiconductor conduction losses.  
         [0053]     A first discharge loop  172  is used in the positive half cycle (PHC) discharge mode wherein switch T 4    183  is turned off and T 3    182  is turned on, while T 1    180  is turned on and T 2    181  is turned off. An approximate equivalent circuit is shown in  FIG. 19 . After the inductor L 1    178  is charged in charge mode, its current i 1  reaches a peak value I′ 0 . During the course of PHC discharge mode, the energy stored in the inductor L 1    178  is transferred both to grid  174  and to capacitor C  184  temporarily, which will be transferred to the grid  174  to support a continuous output during the time when the inductor L 1    178  is being charged again.  
         [0054]     Since the inductor L 1    178  is in the discharge mode, its current i 1  is decreasing. If the change of capacitor voltage, Δν c , is small compared to its absolute value, current i 1  can be regarded dropping linearly. Moreover, current i 1  will drop to zero provided the time duration of PHC discharge mode is long enough. Thereafter it will keep zero until the next occurrence of the charge mode because the diode D 3    176  blocks the current to flow back from the capacitor C  184 .  
         [0055]     Two current conduction modes can be defined here, which we will discuss later. If the time of PHC discharge mode is so short that the inductor L 1    178  starts being charged without its current decreasing to zero when next charge mode comes, the current of energy-storage inductor L 1    178  is continuous, referred to as continuous conduction mode (CCM). On the other hand, if the inductor current drops zero in PHC discharge mode and remains zero for a time defined as the idle time, this will lead to a discontinuous conduction mode (DCM), as illustrated in  FIG. 21 . If the idle time is zero, the condition is known as the critical DCM.  
         [0056]     Generally, in the PHC of ac output, energy is transferred from the dc source  175  to the ac grid  174  through alternate cycles of charge mode and PHC discharge mode.  
         [0057]     A second discharge loop  173  is used in the negative half cycle (NHC) discharge mode which is combined with charge mode to provide NHC ac output when switch T 1    180  is tuned off and T 2    181  is turned on. The approximate equivalent circuit of NHC discharge mode is shown in  FIG. 20 .  
         [0058]     Through flyback operation, the current of the primary side L 1    178  drops to zero suddenly and the current of secondary side L 2    179  reaches to the initial current of primary side L 1    178 , if the inductances and turns of both sides are identical and there is no magnetic leakage.  
         [0059]     The major differences between NHC discharge mode and PHC discharge mode are that the ac grid  174  is in the negative half cycle and the discharging current has an opposite direction. The operation of the NHC discharge mode are similar to that of the PHC discharge mode. The NHC energy is transferred from the dc source  175  to the ac grid  174  through L 1    178 , L 2    179  and C  184  by alternating cycles of charge mode and NHC discharge mode.  
         [0060]     Thus, during the PHC of output, the inverter alternates between the charge mode and the PHC discharge mode. During the NHC of output, the inverter alternates between the charge mode and the NHC discharge mode.  
         [0061]     The inductor current can be controlled in either a discontinuous conduction mode (DCM) or continuous conduction mode (CCM). In both schemes, the purpose is to obtain an output current, i p , so that minimal filtering is required to recover or modulate the sinusoidal current waveform.  
         [0062]     In DCM, at each switching interval, the energy-storage inductor L 1    178  is charged from zero and discharged to zero. The inductor current is discontinuous, and comprises the current through T 1    180  in PHC and the current through T 4    183  in NHC. The unfiltered output current is multiple triangular pulses whose amplitudes are modulated in a sinusoidal way, as illustrated in  FIG. 22 .  
         [0063]     In CCM, the inductor current is controlled to follow a sinusoidal waveform within a small envelope. The unfiltered output current is multiple trapezoidal pulses with amplitudes distributed sinusoidally, as illustrated in  FIG. 23 .  
         [0064]     The output current depends on the low-frequency components of unfiltered output current, and the peak value of unfiltered output current is determined by the inductor peak current. For the DCM scheme, the inductor peak current is dependent on both the maximum charging time and the inductance. The output current is thus affected by the switching frequency and the inductance of energy-storage component to a certain extent. For CCM scheme, the selection of inductor value can be wide because the change of i L  is gradual within the small hysteresis bands. Another advantage of the CCM scheme is the fundamental component of unfiltered output current is larger than that of DCM, provided their unfiltered output currents have same peak values because the area of a trapezoidal pulse is larger than that of triangular pulse.  
         [0065]     Representative component selection will now be discussed. The present invention is well suited to the use of Insulated Gate Bipolar Transistor (IGBT) power switches for T 1    180 , T 2    181 , T 3    182 , and T 4    183 . For small DG energy systems, exemplary IGBT switch parameters are as follows: V dc =300V and P o =1 kW, where the maximum V ce  is about 500V and the average current and peak current are  15 A and  60 A respectively. An example of a typical IGBT is International Rectifier&#39;s IRG4PF50WD. An exemplary power diode D 2    177  and D 3    176  is part 40EPS12.  
         [0066]     The mutually magnetically coupled inductors L 1    178  and L 2    179  can be implemented as a flyback transformer. As is well known in the art, a flyback transformer is a specialized transformer optimized to store magnetic energy.  FIG. 24  illustrates an exemplary flyback transformer  240  for use in the present invention. The flyback transformer core  241  has a high reluctance, typically with an air gap  242 . Current flows in either the primary winding  243  (equivalent to L 1    178 ) or secondary winding  244  (equivalent to L 2    179 ), but not both at the same time, because the energy is stored in the magnetic circuit when the primary side  243  (L 1    178 ) is connected to the source, and transferred to the secondary side  244  (L 2    179  when the primary side is disconnected. The flyback transformer  240  thus comprises two highly-coupled inductors L 1    178  and L 2    179  with equal inductances and number of turns. In an exemplary S 2 B 2  inverter, the flyback transformer is designed to work in DCM with an operating frequency of 9.6 kHz.  
         [0067]     A second embodiment of the present invention is illustrated in  FIG. 26 . This single-stage buck-boost inverter  260  uses only three switching devices  261 ,  262 ,  263 .  
         [0068]     The circuit operation can be divided into four modes. Mode 1 and mode 2 work in positive half cycle, and mode 3 and mode 4 work in negative half cycle.  
         [0069]     Positive half cycle: During mode 1, switching device Q 1   261  is turned on and switching devices Q 2   262  and Q 3   263  are turned off, the coupled inductor L 1   264  will be charged with input voltage  175 ; and during mode 2, switching device Q 1   261  and Q 3   263  are turned off and Q 2   262  is turned on, the energy in L 1   264  will be discharged to the grid through D 2   266 .  
         [0070]     Negative half cycle: During mode 3, switching device Q 1   261  is turned on again (Q 2   262  and Q 3   263  are turned off) and the coupled inductor L 1   264  will be charged; and during mode 4, switching device Q 1   261  and Q 2   262  will turned off and switching device Q 3   263  will turned on, the energy transferred to the coupled inductor L 2   265  from L 1   264 , will be discharged to the grid  174  in reverse direction (compared to the positive half cycle) through D 3   267 .  
         [0071]     In an exemplary embodiment, the inductances of the two coupled inductors L 1   264  and L 2   265  are 0.5 mH, the filter parameters are L=2 mH, and C=25 uF.  
         [0072]     The inverter of the present invention can be controlled using control techniques well known in the art. A technique well suited to the inverter of the present invention is Sinusoidal PWM, also known as subharmonic or suboscillation modulation, and is a carrier-based voltage control method. Its purpose is to synthesize the switch gating signals to the switches in such a way that the output voltage or current waveform is as close to a sinusoid as economically possible.  
         [0073]     Basically, a sine reference wave, serving as modulating signal, is compared with a triangular carrier wave, and the intersection points determine the switching angles and pulse widths as in  FIG. 3 . 1 . The generated switch gating pulses vary proportionally with the modulating signal; in other words, the pulse width is maximum in the middle of each half period and decreases as cosine function towards either side. A variable-frequency variable-amplitude output can be obtained by varying the frequency and amplitude of the modulating signal.  
         [0074]     For a typical SPWM inverter, the spectrum of harmonic frequencies in the output is shifted towards the high frequency and the lower-order harmonics are reduced significantly. Thus, the output filter elements can be smaller to attenuate only higher-order harmonics, which are carrier-frequency-related with modulating-frequency sidebands.  
         [0075]     Sinusoidal PWM control strategy can be used with the S 2 B 2  inverter of the present invention to provide a DCM current scheme. Unlike traditional stand-alone buck inverters, the output current is controlled to be sinusoidal for grid-connected systems. Closed-loop SPWM control provides further robustness and insensitivities to dc and ac variations as well as parametric uncertainties. An exemplary SPWM control circuit of the present invention is shown in  FIG. 25 . Such controllers are well suited to implementation on an integrated circuit, facilitating cost reduction.  
         [0076]     The Single-Stage Buck-Boost (S 2 B 2 ) inverter of the present invention is implemented with four or three power semiconductor switches and can deliver an ac output voltage from a dc input voltage which can be higher or lower than the ac output voltage.  
         [0077]     The embodiment(s) of the invention described above is(are) intended to be exemplary only. The scope of the invention is therefore intended to be limited solely by the scope of the appended claims.