Home >> solar automatic street light control using ldr sensor kit
this paper aims at designing and executing the advanced development in embedded systems for energy saving of lights. nowadays, human has become too busy, and is unable to find time even to switch the lights wherever not necessary. the present system
arduino based auto street light intensity controller of automatic street light control using sensors and solar panel .. and implementation of automatic street light control using sensors and
jun 13, 2018 in this project, we are going to show you how to make a arduino based solar tracker using ldr & servo motor. the solar panel tracker is designed to follow the sun movement so that maximum light intensity hits on the solar panel, thus increasing the power efficiency. we have designed single axis solar tracking system.
description. the circuit diagram present here is that of a street light that automatically switches on when the night falls and turns off when the sun rises.in fact you can this circuit for implementing any type of automatic night light. the circuit uses a light dependent resistor (ldr) to sense the light .when there is light the resistance of ldr will be low.
solar lights. december closing dates 30w solar street light with remote control, automatic day/night switch r 1,399.00 150w solar street light with motion sensor, remote control, automatic day/night switch r 2,299.00 sale. 42 led solar motion sensor wall light r 595.00 r 475.00
sep 11, 2017 these circuits can be used for different automatic light sensing projects including automatic night lamp, automatic street light controller, dark sensor circuits, etc. automatic night lamp circuit using ic 555 and ldr. an automatic night light circuit controls the switching of light by sensing the surrounding light intensity.
sep 29, 2018 here is automatic night light circuit using scr and ldr without relay how automatic sun light sensor switch circuit works the circuits ready working. when the male plug, or for the circuit to the ac line power at 220 volts. the plug socket is connected to
a solar panel that can rotate in two axis either automatically using four light sensors, either manually with the aid of two potentiometers .. solar ups controller/automatic transfer switch temperature sensors with wifi connectivity to control and monitor optimal usage of solar energy. solar water tank regulator. project showcase by
may 06, 2019 a very simple automatic solar light system for illuminating your garden passages can be built using some leds, a rechargeable battery and a small solar panel .. can you provide a solar lamp with ldr circuit details ?, something similar to a ltter of light program .. i want one circuit diagram.and i want automatically on & off the street
day night switch, light sensor switch, automatic light switch, home automation amazon solus 120v automatic dusk to dawn photocell photo control light sensor switch and adapter kit for hardwire outdoor lamp posts (with ezee change plug (black)) street lights, glow sign boards,hoardings or any electrical device ,parking lot
efficiently to turn street lamps on/off. the ldr sensor is the only sensor used in this circuit. the lamps will come on a light system using timer controller is overcome and human intervention is completely eliminated. by this energy con sumption and cost are drastically reduced. the automatic street light control system based on
the portal will have video lectures, tutorials, and quizzes required to build the automated street lighting project. kits are well tested and shipped to your doorstep in 24 hours. use the kit along with the online portal to complete the project at your home itself. you can enroll for free demo to
nov 04, 2012 the drop across ldr is used for switching the transistor. solar led street lighting systems are the advanced versions of this ordinary automatic street light controller. solar table lamp circuit for your home; remote controlled light using 555 timer; circuit diagram of automatic street light using ldr
automatic street light with motion sensor is a very useful and energy efficient technique. relay is used as the automatic switch in this system. it releases the manual work at most up to 100 per cent . as soon as the sunlight goes under the visible
street light that glows on detecting vehicle movement street light that glows on detecting vehicle movement circuit. the proposed circuit is built with atmega microcontroller, ds1307 ic, ldr, lcd, pir sensor, array of led.this circuit is very useful in our day to day life like highways, real time street lights, the parking areas, restaurants and hotels.
aug 21, 2018 in this arduino project, we are going to build an automatic speed controller of dc motor using ldr with arduino. this ldr based arduino project slows down this ldr based arduino project slows down the speed of motor as the intensity of light falling on ldr decreases and vice versa.
dec 10, 2017 automatic street light dimmer using motion sensor will save power by reducing its brightness when no one is passing in the road, pir sensor will detect the motion in the road. the signal from the pir sensor will activate the circuit and increase the brightness, this street lamp circuit not using microcontroller thus we can reduce cost. we can use this simple circuit for outdoor lighting
diy solar rechargeable automatic night light (outside use) in this i will show you how to make a solar rechargeable automatic led night light. i made mine for outside use but hey you could do whatever with yours ) how did this idea come about? first i have been meaning to get some sort of light fi
in this project automatic street light control by pic microcontroller, ldr sense the outside light when sun light is dropped, resistance of ldr decreases and voltage drop across ldr reduce and high voltage is feed to comparator circuit.
smart streetlight using ir sensors srikanth et al  proposed a zigbee based remote control automatic street light system. the system operation of solar street lighting system can be divided into 2 period of time, namely, at 18.00 22.00 hours and 05.00 06.00 hours. the design of a control circuit was experimentally done in this work.
ldr (light depending resistor ) whose resistance is based upon the quantity of the light declining on it along with a few more components. circuit diagram of automatic street light controller switch circuit using relays and ldr is given below. automatic street light controller switch circuit electronicshub.org
auto intensity control of street lights project kit by edgefxkits . furthermore, this project can be developed by incorporating it with the solar panel, as it changes the intensity of the solar into equivalent voltage, and this energy is used for highway lights.
light dependent resistor circuit diagram. in the morning time, this sensor has a low resistance around 100 . thus, the power supply flows through the ldr & ground through the variable resistor and resistor as shown in the above light sensor circuit.
the implementation of street light intensity control using lux meter, traffic sensor and complex sub control machines are in process in the norway. but the power consumption is reduced only by nearly 30 . there also exists a project in progress where in the street light power consumption is reduced using a remote controlled
apr 07, 2016 ldr alone can't do this, you need some other components to make this happen. so let's take a look at a simple circuit using ldr and some other components to explain how it will turn on a light when it is dark. what you see here is a ldr or light d
and the sensor controlled lighting system is solar powered, and the main idea of ldr light sensor circuit, there was a problem with the ldr, it was turning the led on with 5v instead of 12v, thiswas with using 10k .. the project aims to reduce the side effects of the current street lighting system, and find a solution for fuel future
dec 20, 2016 first we study the circuit diagram of the automatic street light .. but the other side of resistor is connected to ldr sensor, ldr sensor is light dependent resistor, it can senses light. if light is present or not. light dependent resistor is not polarity sensitive, so you can connect as you can .. basic introduction of stm32f4 kit the
aug 31, 2012 abstract a street light that automatically switches on when the night falls and turns off when the sun rises. in fact we can use this circuit for implementing any type of automatic night light. the circuit uses a ldr to sense the light . when there is light the resistance of ldr will be low. so the voltage drop across pot r2 will be high.this
this automatic staircase light circuit switch on the staircase lights automatically when someone enters on the stairs and gets off after some time. there are two important components in this circuit, first is pir sensor (passive infrared sensor) and second is relay.
smart street lighting system using iot ms. m. kokilavani, dr. a. malathi research scholar, assistant professor the pir sensor and ldr sensors are used to sense in this diagram street lights control by the sensors. it get the data from object. when vehicles appear to sensor then
automatic streetlight using ldr
this is done by a sensor called light dependent resistor (ldr) which senses the light actually like our eyes. it automatically switches off lights whenever the sunlight comes, visible to our eyes. aim of this seminar is to control the street light using ldr. when the light
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there have been lot of problems in street lights. major problem in some places is every evening a person has to come and switch on the street light and it should be again switched off in morning. yes, this may not be the situation in everywhere but exists in many places. so this problem can be overcome by using
jul 18, 2019 this triggers the alarm and shows a burglary attempt. a lamp is also used to indicate the theft when light falls on the sensor. in future, this project can be developed by using a gsm modem and also a microcontroller. this modem can be interfaced to send an sms to the user in case of burglary. ldr based light intensity control for street lights
light control system using ldr and arduino is a whole new idea in the world of street lights. in the wake of experiencing numerous exploration papers which were found on the lighting system were only based on their working and not on the consumption of energy or electricity the main thought process of doin


|Good sample prep and errors it can avoid
|Example: K2SO4 powder
|Addressing micro absorption
|Addressing preferred orientation
|Addressing particle size
|Sample holders for different needs
|Q & A
is one of the most powerful tool for identifying unknown crystalline phases. Many considerations and implications rest on the XRD results collected in terms of designing better materials or making decisions on your process control. What would your conclusions be if you notice peak shifts or peak broadening in your sample? During our work with numerous clients, we've noticed that the majority of poor XRD patterns is the result of poor XRD sample prepartation. Look forward to practical tips shared by Malvern Panalytical's senior XRD application specialist, Dr Daniel Lee during this webinar.
During this webinar, he shares tips relating to the optimal particle size, how to properly grind powder samples, choice of sample holder for situations like limited powder samples and more. Look out for our upcoming series of webinars on "Better XRD data quality results and interpretation".
Interested to learn more? Join our free series of webinars: "Better XRD data analysis and interpretation for materials characterization":
- Webinar 1: Introduction to powder X-ray diffraction. More info
- Webinar 2: Studying battery cathode materials using X-ray diffraction More info
- Webinar 3: Expand your powder XRD applications for materials characterization research More info
- Webinar 4: Good vs bad XRD patterns: how to improve your phase analysis. More info
- Webinar 5: Better XRD data quality: importance of good sample preparation.
- Webinar 6: Improving your phase search mapping by defining your elemental range: introduction to elemental analysis using X-ray fluorescence. More info
- Webinar 7: Live demo at your desk - latest high performing XRD Benchtop. More info
- Webinar 8: XRD phase quantification tutorial - improve your XRD data analysis. More info
- Webinar 9: XRD data analysis on HighScore Plus version 4.9 - what's new? More info
- Webinar 10: Range of XRD instruments to aid materials characterization research. More info
Dr Daniel Lee, Malvern Panalytical's senior application specialist for XRD, based in South Korea
Dr Lee has a PhD in Materials Science and Engineering. his expertise is on PVD, thin-film analysis, phorphors, solar cells, lithium ion batteries and thin-film battery research. He has been instrumental in providing application support using X-ray diffraction analysis to various industries including electronics, batteries, pharmaceuticals, building materials and more. These include powder diffraction, Rietveld refinement, small angle X-ray scattering, stress and texture research. Prior to joining Malvern Panalytical in 2013, he worked as a senior researcher with GS Nanotech. He was involved in the process engineering process for thin film lithium ion batteries. He also held prior appointments within the R and D team at Telio Solar Korea for CIGS solar cell development.
Who should attend?
• Researchers involved in materials characterization analysis who want to expand their knowledge in X-ray diffraction
• Relevant industries include (but are not limited to) those researching on battery, powder metallurgy, cement, mining and minerals, environmental monitoring, pharmaceuticals and more
• R and D and manufacturing leaders responsible for appropriate analytics selection
• Scientists engaged in method development for new materials or in supporting root cause analysis investigations in support of product manufacturing
How long is this webinar?
40 minutes is the intended speaker time with additional time for addressing queries.


Инструкция на TECO 91-93 (англ.)
Страница 6 из 39
- lower the castor wheel (G fig.4) located on the left side of the machine by turning the two relevant knobs (H fig.4) counter-clockwise.;
- stow the wheel lift in its rest position (locked);
- move the machine using handle B, fig.4.
To lift the machine insert the forks of a pallet truck or lift truck under it so that the c/c centre is roughly aligned with the right side of the cabinet (fig.2). Alternatively you can place the machine on its original pallet, secure it using the original fixing devices and then lift the pallet, again using a pallet truck or lift truck.
Never apply force to the spin shaft when moving the machine.
Take the utmost care when unpacking, assembling, and setting up the machine as described in this heading.
Failure to observe these instructions can lead to damage to the machine and injury to the operator or other persons.
Remove the original packing once you have positioned it as shown on the outside and keep it intact for possible future transport.
Choose the place of installation in strict observance of local regulations regarding safety in the workplace.
If the machine is to be installed outdoors, it must be properly protected by a canopy roof.
Ambient conditions in the place of operation:
- relative humidity from 30% to 95%
- temperature range from 0°C to +55°C
The machine must not be operated in explosive atmospheres.
Place the machine in the chosen position and make sure the surrounding space is commensurate with the minimum clearances indicated in figure 3.


- Designed for Digital Micrometers
Well designed, durable high precision tools that are suitable for various measuring ranges. INSIZE SPC cables are the perfect instrument for assisting digital micrometers in measuring a wide range of components. Easy to use and adjust SPC cable for digital calipers.
Payment & Security
Your payment information is processed securely. We do not store credit card details nor have access to your credit card information.


Soft error rate (SER) testing and measurements of semiconductor circuits with different operating voltages and operating conditions have been performed using the thermal neutron beam at the Radiation Science and Engineering Center (RSEC) at Penn State University. The high neutron flux allows for accelerated testing for SER by increasing reaction rate densities inside the tested device that gives more precision in the experimental data with lower experimental run time. The effect of different operating voltages and operating conditions on INTEL PXA270 processor has been experimentally determined. Experimental results showed that the main failure mechanism was the segmentation faults in the system. Failure response of the system to the operating conditions was in agreement with the general behavior of SERs.
All Science Journal Classification (ASJC) codes
- Analytical Chemistry
- Nuclear Energy and Engineering
- Radiology Nuclear Medicine and imaging
- Public Health, Environmental and Occupational Health
- Health, Toxicology and Mutagenesis


- NTC Thermistors
- NTC Temp Sensors
- Platinum RTD Temp Sensors
- Thermal Protection
- Circuit Protection
Ultra-thin, flexible Thermistor for fast surface temperature sensing especially in tight spots
Truly versatile the Semitec JT ultra-thin NTC Thermistor is widely used across the world in many diverse industries from wearable medical technologies (including blood glucose monitors), heat detectors, CPU/IC monitoring, prismatic and pouch cells, and E-V applications.
Slim but strong, the JT thermistor is only 0.5mm thick at its sensing tip (thinner than a credit card) making it the thinnest flexible NTC available world-wide.
Suitable for both air and surface temperature monitoring- the sensing element is protected/laminated between two polymer layers making it flexible and resilient at the same time. This flexibility allows the JT’s sensor to be placed much closer to the heat source and often in many space-sensitive applications helping to deliver optimum temperature control.
The Semitec JT sensor can literally bend over backwards to solve your temperature sensing problems!
Also, a high performing NTC – the Semitec JT can operate in extreme temperatures from -50°C to +125°C with highly accurate 1% R25 tolerances and resistance values of 10kΩ and 100kΩ. In addition, the JT thermistors have excellent electrical insulation and so can be safely used where they might come into contact with adjacent live components.
Putting it simply, our JT Thermistors outperform other slower, larger inflexible competitors!
Semitec NTC Thermistors form part of an extensive range of thermal components and protection devices available from ATC Semitec. With over 25 years in the temperature sensing and protection devices, our engineers will be able offer expert advice to your new and current projects.
Contact us for comprehensive technical advice, samples and bespoke customer solutions.


Especially the sound analysis of complex objects often reveals the problem that conventional measurements using a single or just a couple of microphones only deliver insufficient results with respect to the tracking of sound sources. The technique used with Acoustic Camera AC100 provides an excellent solution. Simultaneous measurements using a large number of microphones allow the quick and precise location and separation of sound sources.
Typical applications for Acoustic Camera - AC100 are:
- Localization and separation of sound sources to reduce noise emitted by vehicles, machinery, household appliances and electric tools
- Source analysis for acoustic design of wind tunnel models, complex vibrating structures, creaking or rattling structures
- Long-term monitoring of machines and plant
- Acoustic vehicle tests (inside and outside
- Quality assurance
- Predictive machine maintenanc
- Acoustic optimization of product
The handheld acoustic camera AC100 * Compact * mobile * robust * low-cost *
Due to its state-of-the-art microphones in MEMS technology with integrated AD converters and 51.2 kHz sampling rate, together with an integrated measurement data processing electronics in the latest DSP technology, the AC100 is now affordable for wide range of users. The complete integration in the array, with only one USB interface for data and power supply, allows for convenient mobile use with any Windows PC.
The AC100 is a compact and very easy-to-use acoustic camera for standalone applications. Because of its high performance, simple operation and fair price, the AC100 is suitable for a wide range of applications in research and production.
The AC100 is supplied either as pure hardware + Windows driver for custom development or bundled with the BeamformX application software from OptiNav.
Industry Noise Detail
machine bearing at 7128 Hz
Car door opened a bit
car door small leakage
Versions and options
|801961.0||AC100 handheld microphone array|
AC100 BeamformX software
BeamformX uses an advanced new evaluation algorithm, the Robust Functional Beamforming. Compared to conventional beamforming methods, Robust Functional Beamforming works faster, has a higher dynamic range and can resolve more details spatially and temporally. Robust Functional Beamforming has been developed by Robert P. Dougherty, OptiNav Inc USA.
|Dimensions of the array||40 x 40 cm with integrated microphones|
|Microphones||40 MEMS microphones with 24 Bit ADC|
|Sampling rate||51.2 kHz, decimation selectable|
|Anti-aliasing filter||yes, integrated digital filter|
|Frequency range||200 Hz to 24 kHz|
|Maximum sound pressure level||112 dB|
|Dynamic range||20 dB (5 kHz to 20 kHz)|
|Working distance||0.5 m to infinity|
|Front-end||signal processor integrated into array
data and power connection via USB 2.0
|Optical camera||integrated 5 megapixel camera|
|VESA holder||10 x 10 cm with tripod thread 3/8"|
|Optional expansion||4 digital I / O (2x In, 2x Out)|
|Temperature range||−10°C to +45°C|
|PC System Requirements||Win7 or higher 64 Bit, 8 GB RAM|
|Software||API or BeamformX (Optinav)|


Posts with the tag: optical phase conjugation
In the last few years, several papers have looked at how it might be possible to use a multimode fibre as an ultra-narrow endoscope (see this post and this post for a bit of background). The most common approach is to use a spatial light modulator to shape the wavefront entering the fibre. If this is done in precisely the right way, interference between light coupled into the different modes of the fibre will result in a focused spot at the far end. By adjusting the input wavefront it’s then possible to scan the spot in two dimensions, allowing point-by-point imaging. Of course, we need to know what wavefronts to use, making it necessary to perform a calibration which requires access to the far end of the fibre. Unfortunately, this calibration is highly dependent on the configuration of the fibre – if the fibre is bent then the calibration changes. This means the technique is only applicable to rigid probes, greatly limiting the scope of potential applications. Now, in a paper published in Nature Photonics, Tomáš Čižmár and colleagues from the University of Dundee have suggested a possible solution to this problem.
Optical imaging into thick tissue is generally limited to a penetration depth of only a few millimetres. Beyond this depth, almost every photon is scattered multiple times before reaching the detector, meaning that we no longer have much of an idea where it originated from. It’s possible to sidestep this limit if we can somehow differentiate photons that have come from a single point in the sample (for example by using a guide star or ultrasound to modulate the optical signal), and if we measure the wavefront that emerges from the sample. Then, using a technique called optical phase conjugation, a reversed wavefront can be sent back into the sample. Since this wavefront will undergo the opposite series of scattering events to the outcoming beam, it will be focused back to the original point. This ‘time-reversal’ technique can then allow us to image (by scanning the beam and collecting all the returning light or fluorescence) or to deliver various kinds of laser-therapy. However, the difficulty is that the scattering depends on the positions of all the scattering particles in the tissue; if there is movement then wavefront measurement becomes invalid. An international group of researchers have recently shown that the time-constant of the optical phase conjugation is linked to another property of scattering media – speckle correlation. The paper reports measurements of these time constants, giving an indication of how often the wavefront measurement would be needed.


M-2299 Adapter Panel
The M-2299 is an adapter panel which, when combined with the M-2001 Series Digital Tapchanger Control, provides convenient direct replacement of Toshiba TB-R800 Regulators. The M-2299 mounts into the control cabinet using the same two hinge pins which must be saved from the original regulator.
- Connects easily to the M-2001 Series Digital Tapchanger Control using mounting screws and 24-pin connector
- Provides direct mechanical replacement of the existing regulator control
- Provides built-in CT shorting protection when the M-2001 Series Digital Tapchanger Control is removed


ARGO Real Time Compound Inspection System
The ARGO equipment is a quality control device for compound distribution systems on circular supports. It is a simple and irremissible tool for the quality control on the product and of the production phases.
The reading system is composed of a double optical fiber protected by a sheathing which is particularly resistant against external agents and mechanical stress. The fiber ends with a sensor that works on the color contrast. The sensor is encapsulated in a stainless-steel case, which is particularly resistant against shocks. A focusing system with micrometrical regulation is situated on the final part of the case. For the light source, which is obtained by means of led diodes, there is no loss of efficiency or aging. The best reading distance is of 12~15 mm between the inspection surface and the final edge of the reading device.
The light beam of the sensor is oriented by fixing the sensor with a special support, which is different according to the various applications. The fixing and regulation of these supports is very easy. With the support the sensor can be orientated for a partial reading of the mastic area under the curler. All placements and regulations are easily performed and there is no need of intervention of specialized personnel. The design of the support has been particularly accurate and allows the removal and the replacing of the sensor for the periodical cleaning. When performing this operation there is no need of further setting up. The initial set-up is very easy, because it is immediately possible to check the focusing looking at the light beam on the surface that should be inspected.
Finally, the sensor does not need any maintenance intervention, except for the cleaning of its final part, which can be exposed to accidental mastic splashes.
By means of the sensor sensitivity, the mechanics, the optical system and the possible regulations it is possible to detect any smallest imperfection concerning the distribution of the mastic film.
The information read by means of the sensor is then sent to the control system for processing.
Height 104 mm
Width 266 mm
Depth 200 mm
Voltage 220Vac ± 10%
Frequency 50/60 Hz
Power 12Vdc ± 1%
Typ. tension 12Vdc
Max. tension 15Vdc
Alphanumerical Lcd with backlight
Matrix LCD membrane
max. 550 pieces/min.
max. 3.000 revolutions/min.
max. 10.000 pieces
Interruption diameter 0.5mm
Automatic Basic Compound Mixing & Pumping StationThe GEM Basic Automatic Compound & Mixing station allows easy & simple compound management.The system is designed to keep the compound in optimal working condition.The station is...
Train SimulatorGEM Compound Solution also projects and build other types of Solution for its customers. Fox Example we have designed a Train Simulator SIMURAIL also reproduces the instrument panel and instructor's console. The instrument panel...
Electromagnetic RM07 Compound GunThe new electromagnetic RM07 gun has the particularity that it runs solely with electrical current (generated by the GEM Gemini or Gemini ROSSO Control Unit) without the need of compressed air (MKIII). The...


MACHEREY-NAGEL AQUADUR test strips for determining the water hardness
Quick and easy determination of water hardness
AQUADUR test strips are for the quick and easy determination of water hardness. The convenient dip & read procedure and the clear colour changes from green to red ensure reliable results within seconds.
The water hardness depends mainly on the amount of calcium and magnesium salts within the water. The total sum of these salts determines the actual water hardness. In the USA, water hardness is expressed in ppm (mg/l) CaCO3
Generally, water is often simply classified as "soft" or "hard" water. By and large, the following values apply to these terms:
- below 50 ppm CaCO3 very soft water
- 50 - 120 ppm CaCO3 soft water
- 120 - 240 ppm CaCO3 medium hard water
- 240 - 360 ppm CaCO3 hard water
- above 360 ppm CaCO3 very hard water
Water hardness plays an important role in everyday life and a wide variety of industries and applications. Hard water can even adversely affect taste and appearance of food. Hard water inhibits the lathering of soap, so more detergent has to be used in hard water in order to achieve the same cleaning results as in soft water. Additionally, hard water may cause calcification and can thus damage machinery, appliances, pipes or heating units.
In many industries, water hardness is an important parameter in process control and has to be monitored constantly. Some examples of industries include the textile industry, dyeing factories, large scale laundries, water works and fish-farming and aquaculture.
AQUADUR water hardness test strips come in a wide variety of different gradations and packaging options for your individual needs. Single sealed test strips are for example available for promotion activities to inform customers about the necessity of water softeners.
Since we completely control the production process of our test strips in-house, we can cater to very specific or unique requests.
AQUADUR water hardness test strips are the ideal choice to quickly, easily and reliably determine water hardness.MACHEREY-NAGEL AQUADUR Sensitive test strips for determining the water hardness
Highly sensitive determination of low-range water hardness
AQUAUDR Sensitive water hardness test strips include all the same benefits as the classic AQUADUR test strips, but feature a special, very low measuring range. The test strips are the perfect tool whenever very low amounts of water hardness have to be safely detected.
The test strips are for example successfully used in dialysis centers to check feed water, so that the sensitive reverse osmosis units are not damaged by excess water hardness.
AQUADUR Sensitive water hardness test strips are also CE marked according to the directive for medical products 93/42/EWG.


FOR A BETTER BEAM QUALITY
Our high quality diffraction gratings are perfectly suited for beam combining of high performance lasers.
Based on our new grating fabrication process, we can produce gratings up to 135 mm in diameter without any stitching errors. This means that you get large gratings with very good homogeneity and minimal wave front distortion.
High Efficiency Transmission Grating for:


Alexander Gaeta's heavy research into light technology promises to lead to speedy microchips that use streams of photons barreling down microscopic waveguides instead of electricity flowing through minute wires.
Gaeta, a professor of applied and engineering physics at Cornell University, and Michal Lipson, a Cornell electrical and computer engineering assistant professor, are leading a research team that's investigating the potential of photonic silicon chips. The team recently created a chip-based broadband light amplifier, a potentially key component in future photonic circuits. Academic labs worldwide are anxiously pursuing photonic silicon research, aiming to create devices that mix photonics and electronics on the same inexpensively manufactured chip (see photo).
The new broadband light amp chip utilizes a technique known as four-wave mixing in which the signal destined for amplification is "pumped" by another light source inside a waveguide measuring only 300 by 550 nanometers. The photons in the pump and signal beams are tightly confined within the waveguides, allowing for a transfer of energy between the two beams. The chip's waveguides are silicon channels surrounded by silicon dioxide.
Since silicon doesn't emit light very well, Gaeta and his researchers turned to an obscure phenomenon known as the Raman effect to amplify light within the chip. The Raman effect relies on the fact that when light is transmitted through matter, part of the light is scattered in random directions. A small amount of the scattered light has frequencies that are removed from the frequency of the incident beam by amounts identical to the vibration frequencies of the scattering system, explains Gaeta. This light is called Raman scattering. If the initial beam is highly intense and monochromatic, a threshold can be reached beyond which light at Raman frequencies is amplified, builds up strongly and begins showing the characteristics of a stimulated emission.
Previous attempts at using the Raman effect for silicon-based optical amplification were hampered by that fact that signals could only be produced at a single wavelength. Gaeta's team, however, was able to generate signals across several wavelengths. "We have demonstrated that a particular effect can amplify almost 30 wavelength channels at once," says Gaeta. The technique can also be used to create a duplicate signal at a different wavelength, allowing the technology to easily convert a signal from one wavelength to another. Although other researchers have already created four-wave mixing amplifiers using optical fibers, these devices are tens of meters long, hardly small enough for chip-based applications.
The research team tested its invention with infrared light generated near 1,555 nm, the wavelength used in most fiber-optic communications. At 28 nm, the researchers were able to achieve amplification over a range of 1,512 nm to 1,535 nm. Longer wavelengths provided even greater amplification in a range from 1,525 to 1,540 nm. Gaeta predicts that much better performance could be obtained by refining the process. "We're still kind of feeling our way around," he says.
Gaeta also believes that four-wave mixing applications that have already been demonstrated in optical fibers will now be possible in silicon. These applications include all-optical switching, optical signal regeneration and optical sources for quantum computing. He predicts that hybrid photonic/electronic circuits will first be used in data repeaters and routers, where several different wavelengths are sent over a single fiber at the same time.
Innovations like the chip-based light amp will prove to be pivotal in moving photonics from long-haul telecom networks to computer backplanes, says Gaeta. "As the need to transmit data between processors outstrips the capabilities of pure electronic technology, photonics will step in as the solution," he says.


The HillSpeed Digital Speedometers provide an additional or alternative vehicle speed display. They can be used in a range of situations where an accurate, easy to read, prominent display of speed is required and where warning of exceeding a maximum speed is either necessary or desirable. These can include:
- general motoring where speed limits apply.
- industrial applications where there are OH&S requirements.
- driver training and examinations.
- where the existing vehicle speed display requires replacement.
- where there is no existing speed display.
- situations where constant vehicle speed is required.
- where vehicle modifications such as different size wheel rims mean that the existing speed display is not accurate.
Both models of the HillSpeed Digital Speedometer accurately monitor speed and are designed to be mounted on the top of the dash of any vehicle, within the driver’s field of vision.
They have a large, easy to read display in all light conditions, with automatic brightness adjustment according to the light conditions. They are easily and safely read at a glance with no need for the driver to take their eyes off the road. The driver is aware of their speed at all times, allowing them to drive safely and avoid speeding fines.
The in-built overspeed alarm can be set to any desired speed and quickly and easily programmed to give any combination of the in-built audible and visual alarms when that speed is exceeded.
Both models have plug and play operation, allowing them to be simply and easily installed and operating immediately after installation.
Click here to purchase a HUD 13 or HUD 09 digital speedometer.


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17 September 2011
LabVIEW is a graphical programming environment used by millions of engineers and scientists to develop sophisticated measurement, test, and control systems using intuitive graphical icons and wires that resemble a flowchart. It offers unrivaled integration with thousands of hardware devices and provides hundreds of built-in libraries for advanced analysis and data visualization – all for creating virtual instrumentation. The LabVIEW platform is scalable across multiple targets and OSs, and, since its introduction in 1986, it has become an industry leader.
16 October 2010
10 October 2010


CEREBRO: an electric Contrast medium for computationally intensive Electroencephalographies for high REsolution BRain imaging withOut skull trepanation
Project identification number:
Imaging the brain activity is fundamentally important for many brain-related scientific disciplines. Among the non-invasive neuroimaging strategies, Electroencephalography (EEG) from scalp potentials is one of the primary. In EEG the neuroninduced electric potential is measured by using electrodes on the patient’s scalp. The skull however, highly resistive, shields EEG recordings limiting the spatial resolution. The standard way to avoid skull shielding effects is to invasively implant EEG electrodes under the skull (ECoG) or in the brain cortex (StereoEEG), in both cases after trepanning the patient’s skull. Scalp EEGs are noninvasive but lack spatial imaging accuracy. ECoG and StereoEEG are highly accurate but require skull trepanation and they image only a limited part of the brain. There is the need for increasing the resolution of scalp EEG providing the same level of accuracy of invasive EEGs. This will be the grand challenge which CEREBRO will achieve by conceiving the first ever existing EEG contrast medium, able to provide imaging of the entire brain and in a non-invasive way.
- Francesco Paolo Andriulli. (Responsabile Scientifico)
- POLITECNICO DI TORINO - Coordinator
- Ecole Polytechnique Fédérale de Lausanne
- G.TEC MEDICAL ENGINEERING GMBH
- INSTITUT MINES-TELECOM
- UNIVERSITE DE BRETAGNE OCCIDENTALE
Sustainable Development Goals
|Total cost:||€ 2,498,688.00|
|Total contribution:||€ 2,498,688.00|
|PoliTo total cost:||€ 1,380,526.75|
|PoliTo contribution:||€ 1,380,526.75|
Project website link.
Poliflash article of 25/10/2022.


Fire Sensor TF-001 is a smart fire detector which can be voice activated wirelessly. It can detect and help blow away harmful gases. Once the environment is filled with smoke, it not only sends voice alert, but also provides laser guides to point to evacuation direction.
- Installed using a simple adhesive back
- Compatible with lightings
- Powered by 3V Lithium battery
- One year warranty
- Dimension :122 x 38(mm)


A radically new design makes SCOT the lightest available CB Time Interval Meter.
High sampling frequency of up-to 10 KHz offers high timing resolution of 0.1 mS.
Mains as well as rechargeable Li-ion battery operated. In built battery monitor and cut off circuit for protection against battery deep discharge.
In built real time clock, memory and printer*. Test time stamping on each record enables time references for future use.
PC Downloading & Analysis including 'Trend Analysis' is possible with «CData SCOT» Software supplied with kit.
Instrument debug options like contact status check, command check, memory check, printer check*, PC downloading check are provided.
Simple operation. Tests can be conducted even by technicians.
Supplied with rugged test lead set, suitable for testing EHV CBs.
Enclosed in ultra rugged IP67 class hard molded case.The instrument is supplied with CData software CD, Calibration Certificate, and Test leads suitable for testing of EHV class circuit breakers.
Type Tested as per IEC 60068, IEC 61326 & IEC 61010-1. CE marked


Let’s understand working of refrigerant based dehumidifier which also know as condensation dehumidifier.
Condensation dehumidifier equipped with compressor, evaporator coil, condensation coil and tank. lets understand process
– Moist air drawn into unit by fan which passes over
cold surface of condenser coil hence moisture in
air get condense into water.
– Condensate get accumulated in tank.
– Cold air then passed over evaporator coil and used
for heat recovery system.
– Dry air then pushed in room with 2° Celsius warmer.
– Sensor installed shuts down unit if tank gets full.
– Humidity sensor continuously read humidity level
of air entering in unit. When room humidity reaches
to set level then unit get switched off automatically.
CtrlTech dehumidifier UAE
PO 22271, SAIF Zone, Sharjah, UAE.
T: +971 50 1537113 | +971 55 2338912
Email : Sales@ctrltechnologies.com


The six-axis MITSUBISHI MELFA RV-4F-D industrial robot trainer kit is used for manufacturing with high speed and high precision performance and combining intelligent technology, it has easy connectivity with Mitsubishi’s PLCs and FA Equipments. Vertical multiple-joint type with standard arm type robot used for multiple application like shorting and packaging. The system has inbuilt PLC. RT tool box software which is useful for programming and simulation. Students can design different type of applications and simulate with the help of software simulation module and test on the available hardware.


Shackle pin load cell
When it comes to measuring loads in weighing, rigging or lifting applications, the most common solutions are either using a shackle with an integrated instrumented load pin, or a custom-made load pin sensor that can fit into the existing shackle, replacing existing shear pins, clevis pins, shear pins or any other type of pin joint. This labor-intensive configuration is resource-intensive, as designing and manufacturing a custom pin load cell is a time-consuming process, especially since the designer must ensure that the load applied to the load pin is centered or evenly distributed in the clevis pin.
FUTEK offers a simpler, yet equally effective solution. Instead of replacing or customizing components, a commercially available "wafer" type load cell can be integrated into the existing load block configuration with minimal interaction with the lift and load block components. A high capacity pancake load cell with axially mounted rod end bearings is connected to the load block and lifting hook using existing shackles and load pins, leaving the existing lifting elements largely untouched.
FUTEK's LCF series of load cells are suitable for capacities up to 100,000 lbs and can withstand up to 150% overload with IP protection up to IP65 (higher seals are available upon request). LCF Series load cells are built to withstand the demands of harsh environments and weather conditions and allow for accurate load monitoring and recording at any attachment point.
FUTEK LCF load cells have fatigue resistance, a feature that is particularly important for fixed and long-term applications. The load cell's ability to withstand successive load cycles for long periods of time without risk of failure or damage to the load cell body (bending) or strain gauge element, saves time and resources while providing peace of mind to the operator.
Typical Applications :
- Weighing cranes under hooks
- Show and theater assembly
- Aircraft manufacturing and repair
- Lifting and lowering of loads
- Testing and certification of structural loads
- Shipping and receiving weighing
- Crane and hoist calibration and proof load testing
- Manufacturing and heavy construction
Benefits of LCF Series Load Cells :
- Simple installation and operation
- Fatigue resistant load cell
- High accuracy with ±0.10% non-linearity.
- Eccentric loading - capable of withstanding high external loads
- Stainless steel or anodized aluminum flexure
- Ideal for use in rugged or harsh environments
- Environmentally sealed to IP65 (higher sealing available upon request)
- Available up to 100,000 lbs (higher ratings available upon request)
How it works :
- FUTEK's LCF "wafer" type load cells are installed between the load block and the lifting hook to measure the crane's lifting load.
- As the load is lifted by the crane or other lifting system, the strain gauge load cell captures the weight of the lifted load in real time. At any time, as the load increases, the operator can reverse the lift in case of overload. A load indicator or digital display is installed in the operator's cab for online monitoring.
- The IPM650 intelligent digital display is combined with sensors for real-time recording, locking and alarming.
- These systems are also available with wireless/Bluetooth solutions for improved versatility and logistics.
Products used :
LCF series high capacity pancake load cell combined with IPM650 intelligent digital display ;


Circuit arrangement for frequency analysis of received signals which lie in a wide overall frequency band
A circuit arrangement for a frequency-dependent analysis of received signals which lie in a wide overall frequency band utilizes a filter bank which cooperates with an evaluation circuit. The filter bank has frequency-dependent band filters connected thereto and comprises an evaluation frequency range which exhibits only a few small band width in relation to the overall frequency band. The received signals are processed by means of a filter circuit having adjoining pass ranges, the signals being modulated by a number of frequency modulators into an evaluation frequency range and an addition of the individual heterodyning frequency values is performed. The frequency-dependent successive partial frequency bands of the received signals, after conversion in the filter circuit, are respectively connected to another of two transmission channels in such a manner that the first transmission channel only transmits the even numbered partial frequency bands in the frequency-dependent succession, and the second transmission channel only transmits the uneven numbered partial frequency bands in the frequency-dependent succession. The transmission channels are coupled to one of two respective equally structured filter banks and all outputs of the filter circuit, engaged by means of received signals, supply the output signal which characterizes engagement. Each received signal, if it falls in only one pass range, passes only one overlapping-free transmission path; if it falls into two adjoining pass ranges, it passes two overlapping-free transmission paths and thereby results in unequivocal frequency values for evaluation.
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1. Field of the Invention
This invention relates to a circuit arrangement for the frequency-dependent analysis of received signals which lie in a wide overall frequency band utilizing a filter bank which cooperates with an evaluation circuit, and more particularly to such an arrangement in which the filter bank has frequency-dependent band pass filters and comprises an evaluation frequency range which exhibits only a few small bandwidth in relation to the overall frequency band, and wherein the received signals are processed by a filter circuit having adjoining pass ranges so that the receiving signals are modulated into the evaluation frequency range by means of a number of frequency modulators, and wherein the individual heterodyning frequency values are added.
2. Description of the Prior Art
A radar apparatus is known from the publication "Proceedings of the IRE", Vol. 49, 1961, No. 9, pp. 1417-1423, in which a frequency analysis is undertaken in order to determine the Doppler frequency. The appertaining circuit arrangement operates with a first filter bank, roughly subdividing the practicable frequency range and consisting of a number of band pass filters connected in parallel. On the basis of the rough information thus obtained, that oscillator is then selected from a number of oscillators, by means of a corresponding selection circuit, which permits bringing the received signal into the pass range of a second filter bank which comprises a number of very narrow-banded band pass filters. As a certain time passes for the first frequency analysis in the first filter bank, roughly subdivided, the received signal, in particular the reflected echo pulse, is stored for such a time until the necessary preliminary decisions are made, i.e. the selection of the respective low frequency oscillator. Not until then is a received signal passed on to the finely subdivided second filter bank for a further evaluation.
This known circuit arrangement has the disadvantage that its construction is very expensive. Moreover, it is then no longer suited if a very wide frequency band is to be monitored or is to be examined for the presence of signals.
A difficulty generally occurring in known circuit arrangements which serve for a frequency-dependent analysis of received signals resides in the fact that signals which fall into the transition range, e.g. a "gray range", between two successive pass ranges are either lost in the evaluation due to diminished sensitivity, or provide indication errors. This cannot be completely prevented, no matter how steep the filter response flanks, whereby the disadvantage exists that such filters are complicated and expensive.SUMMARY OF THE INVENTION
The object of the present invention is to improve a circuit arrangement of the type initially mentioned, with the least possible expense in regard to filter properties, in such a manner that the analysis of very wide frequency bands is possible, and where the difficulties which result between the pass ranges of the individual partial frequency bands by means of the "gray ranges" are also avoided.
The above object is achieved in that the frequency-dependent sequential partial frequency bands in the filter circuit are respectively conveyed to a second or two transmission channels in such a manner that the first transmission channel transmits only the even numbered partial frequency bands in the frequency-dependent succession, and the second transmission channel only transmits the uneven numbered partial frequency bands. The transmission channels are respectively conveyed to one of two equally constructed filter banks (filter bank A, filter bank B in FIG. 4) and all outputs of the filter circuit, busy with received signals, provide an output signal for the evaluation characterizing the engagement thereof by the received signal. Each received signal, if it falls in only one pass range, passes only one overlapping-free transmission path; however, if it falls into adjoining pass ranges, it passes two overlapping-free transmission paths and thereby results in unequivocal frequency values when evaluated.
In this manner it is safeguarded that an unequivocal statement concerning the magnitude of the received frequency is obtained for received signals lying in "gray ranges", and that the sensitivity is also retained. The expense for the filter circuits can be maintained relatively low in that no expensive filters having steep characteristics are required.
In accordance with an advantageous development of the invention, the filters are structured such that the individual adjacent pass ranges overlap one another in regard to their transmission loss in such a manner that the received signals lying in the "gray range" and falling into pass ranges are attenuated as little as possible.BRIEF DESCRIPTION OF THE DRAWINGS
Other objects, features and advantages of the invention, its organization, construction and operation will be best understood from the following detailed description, taken in conjunction with the accompanying drawings, on which:
FIG. 1 is a schematic illustration of an input filter group I;
FIG. 2 is a schematic illustration of a pair of subsequent filter groups IIA and IIB;
FIG. 3 is a schematic illustration of a pair of subsequent filter groups IIIA and IIIB, in accordance with FIG. 2;
FIG. 4 is a schematic illustration of two equally constructed filter banks A and B which follow the structure of FIG. 3;
FIG. 5 is a schematic illustration of the logical linkages between the first filter bank A and the preceding filter groups;
FIG. 6 is a schematic illustration of the logical linkages between the second filter bank B and the preceding filter groups; and
FIG. 7 is a graphic illustration which exemplifies the pass ranges of a filter.DESCRIPTION OF THE PREFERRED EMBODIMENTS
FIG. 1 illustrates the first filter group which is located at the input side of the apparatus and which is collectively referenced as the filter group I. Several antennae A1-A4 are provided for receiving signals to be analyzed, the antennae being connected to the filter group I by way of a coupler AK. This antennae arrangement is necessary because a single antenna is not sufficient for the operation in correspondingly wide receiving frequency bands. In the present exemplary embodiment it is assumed that the receiving frequency range to be analyzed lies between 0.3 and 20.3 GHz. A first filter circuit W1 is connected at the output of the antennae coupler AK, the first filter circuit dividing the receiving frequency range into two frequency bands. The frequencies between 0.3 and 16.3 KHz are fed to a filter circuit W2 which is connected to the output of the filter circuit W1 as a residual frequency band for additional analysis. A specific first partial frequency band, having a width of 4 GHz in the present example, is taken from the second output of the filter circuit W1 so that frequencies between 16.3 and 20.3 GHz may be obtained. The respective limiting values of this and all subsequent partial frequency bands include larger "gray zones", and is explained later on in connection with FIG. 7, which zones are provided by means of the overlapping of the pass ranges of the filters. An output voltage U1 is tapped from the filter circuit and corresponds with the voltage value in the first partial frequency band of 16.3 through 20.3 GHz, which voltage is fed to a mixer M1. A frequency f01=15 GHz is fed to the mixer M1 so that the intermediate frequency band obtained therefrom lies between 1.3 and 5.3 GHz in the frequency range. The first intermediate frequency band is fed to a transmission line referenced channel A.
The filter circuit W2 branches off a second partial frequency band which lies between 12.3 and 16.3 GHz, from the frequency band of 0.3 through 16.3 GHz which is applied to the input of the filter circuit W2. An output voltage U2 is obtained from the second partial frequency band by means of a coupler K2. These signals are fed to a mixer M2 which has an heterodyning frequency selected as f02=11 GHz. The output signals thus obtained and lying in the joint intermediate frequency band of 1.3 through 5.3 GHz reach a second transmission line referenced channel B. The residual frequency band between 0.3 and 12.3 GHz which is available at the upper output of the filter circuit W2 is fed to a subsequent filter circuit W3.
The filter circuit W3 separates a residual frequency band of 0.3 through 8.3 GHz and supplies the frequency range of 8.3 through 12.3 GHz as a third partial frequency band. The signals in the latter band are fed to a mixer M3 having an heterodyning frequency selected at f03=7 GHz. The intermediate frequency band lying between 1.3 and 5.3 GHz is fed to the channel A.
The signals of the upper output of the filter circuit W3 are fed to a following filter circuit W4. The filter circuit W4 supplies the fourth partial frequency band between 4.3 and 8.3 GHz at its lower output, and forms a voltage U4N by means of a coupler K4N. One could obtain a conversion into the intermediate frequency band of 1.3 through 5.3 GHz, required for the channels A and B, by a correspondingly low heterodyning frequency of 3 GHz; however, it is more expedient to undertake the conversion in such a manner that one can operate with relatively high heterodyning frequencies. For this reason, and in order to avoid undesirable mixing products, the heterodyning frequency of a mixer M4N, connected to the output of the filter circuit W4, is selected at the frequency f04=8 GHz. The sum frequency of the signals is filtered out by a band pass filter BP4N and the signals thus obtained are fed to the mixer M2. The output signals of the mixer M2 also lie in the intermediate frequency band of 1.3 through 5.3 GHz. These signals are fed to the channel B. The "resulting heterodyning frequency" from the individual heterodyning frequencies f04 and f02 thus amounts to 3 GHz because the partial frequency band of 4.3 through 8.3 GHz is modulated at 1.3-5.3 GHz in the end result.
The upper output of the filter circuit W4 supplies a residual frequency band between 0.3 and 4.3 GHz and thus simultaneously represents a fifth partial frequency band. An output voltage U4T is obtained by way of a coupler K4T, and after conversion output signals having an heterodyning frequency range f04=8 GHz are obtained from the mixer M4T, these signals having the upper side band filtered out by means of a band pass filter BP4T. These signals are additionally fed to the mixer M3 so that output signals result which also lie between 1.3 and 5.3 GHz and these output signals are fed to the channel A. Double conversion is used to avoid undesirable mixing products. The resulting heterodyning frequency amounts to 1 GHz, wherein the sum frequency--in contrast to the preceding conversions--and not the differential frequency, is fed to the channels A and B.
It should be pointed out that the filter circuits provided at the outputs of the mixers M1, M2 and M3 prevent the transmission of the sum frequencies resulting from mixing, are not illustrated in FIG. 1, nor in subsequent drawings. The connecting points, respectively marked as heavy dots, are to indicate that non-interacting split connections are to be provided at these points, for example, circulators or hybrid circuits, which prevent a reciprocal influencing of the signals which are coupled in. If the different frequency values are to be simultaneously present in the evaluation circuits, delay devices must be provided, in the well known manner, which compensate for the different processing times which have occurred, for example, between the first and the last partial frequency band after their respective conversions into the common first intermediate frequency band (i.e. into the channels A and B).
If the frequency bands of the antennae A1-A4 fit into the frequency schedule of the filter circuits, the antennae can also be connected directly to the respective filter circuit. If, for example, the antenna A4 has a receiving frequency range of 0.3 through 8.3 GHz, its output signals can be fed directly into the input of the filter circuit W4.
Viewing the circuit arrangement in accordance with FIG. 1 results, in summary, that from the receiving frequency range of 0.3 through 20.3 GHz receives by the antennae A1-A4, respectively, subsequent partial frequency bands of 4 MHz are filtered out and are brought into a single common intermediate frequency band of 1.3 through 5.3 GHz by means of conversion. The intermediate frequency band of 1.3 through 5.3 GHz, respectively obtained from the frequency-dependent successive arrangement of the partial frequency bands is distributed to the channels A and B in such a manner that the channel A obtains the signals of the intermediate frequency band formed from the partial frequency bands in uneven-numbered sequence, and that the channel B obtains the signals of the intermediate frequency bands formed from the partial frequency bands in even-numbered sequence. The first partial frequency band (16.3-20.3 GHz), the third partial frequency band (8.3-12.3 GHz) and the fifth partial frequency band (0.3-4.3 GHz) are fed to the channel A, whereas the channel B obtains the second partial frequency band (12.3- 16.3 GHz) and the fourth partial frequency band (4.3-8.3 GHz). Therefore, the received frequencies are transmitted in the channel A, which originally were between 16.3 and 20.3 GHz, between 8.3 and 12.3 GHz, and between 0.3 and 4.3 GHz. In contrast thereto, the channel B contains the original received frequencies of between 12.3 and 16.3 GHz, and between 4.3 and 8.3 GHz. However, the use of the common intermediate frequency band in the channels A and B, there cannot be a differentiation as to which original frequency range the signals may have been received. This differentiation, i.e. the indication, which partial frequency band has supplied the respective output signals, is obtained by the output voltage U1, U2, U3, U4N and U4T which are called upon, in a manner to be described later on, for evaluation as additional information. The division into individual partial frequency bands can proceed at random per se, whereby the view points of undesired products of mixing and the relative bandwidths (for filter evaluation) are to be taken into consideration.
The attenuation of the pass ranges, in dependence upon frequency, is plotted in FIG. 7. The numerical values, which occur in the filters W1 and W2 in FIG. 1, are referred to in detail. The first partial frequency band extending from 16.3 through 20.3 GHz is fed to the channel A. The second partial frequency band between 16.3 and 12.3 GHz is fed to the channel B, and the third partial frequency band lying between 8.3 and 12.3 GHz is again fed to the channel A. If the filter circuits W1 through Wn of the various figures are not expensively constructed, relatively flat ascents in the area of the filter response flanks result. This is illustrated in FIG. 7 by means of the drawn lines. According to this illustration, greater overlappings occur in the area of the limiting frequencies in the different partial frequency bands. These "gray zones" are made noticeable by the overlapping arrows in FIG. 7.
It is indeed possible, by means of the dash-dot lines, to reduce these gray zones by means of a greater filter expense. However, it is not possible to remove the gray zones completely. Moreover, considerable disadvantages result in that with steeper attenuation curves at the flanks of the filter characteristic, the total circuit arrangement becomes very insensitive in the transition range, for example, at 16.3 GHz. A relatively weak signal of 16.3 GHz is very heavily attenuated by the filters corresponding to the dash-dot lines and is therefore, perhaps not admitted and is thus lost for further evaluation. The overall circuit in this range is very insensitive. The filters corresponding to the solid lines, on the other hand, hardly attenuate the signals even in the range of 16.3 GHz and thus bring about an essentially more advantageous sensitivity of the overall arrangement. Accordingly, the various filter circuits in the range of the limit frequencies between the individual partial frequency bands are constructed such that their attenuation a has not yet risen about a tolerable value. Allowances can readily be made for the gray zones which occur because a differentiation is obtained in that one time the signal is fed to the channel A and the next time the signal may be fed to the channel B. When a signal is received at the frequency of 16.3 GHz, indeed not only the first partial frequency band, but also the second partial frequency band, is engaged. In a further evaluation, as described later on, a double indication of the same frequency value in the end result is obtained if the received signal lies in the gray range.
The output signals of the channels A and B are fed to additional filter groups, as exemplified in FIG. 1, these filter groups being more precisely illustrated in FIG. 2. The channel A is fed to a filter group IIA by way of a terminal 2a. Thus, the frequency range to be processed lies between 1.3 and 5.3 GHz. As in FIG. 1, there is a branching of additional, frequency-dependent narrow partial frequency bands by way of a plurality of filter circuits W5A-W7A which are connected in a chain. The width of the new partial frequency bands in this circuit is only selected at 1 GHz so that the frequencies between 4.3 and 5.3 GHz are present at the lower output of the filter circuit W5A (first partial frequency band of the filter group IIA). An output voltage U5A is produced by way of a coupler K5A. The signals thus obtained are fed to a mixer M5 which has an heterodyning frequency selected at f05=4 GHz. Thereby, signals are obtained whose differential frequency lies between 0.3 and 1.3 GHz (the second intermediate frequency band). These signals are fed to a channel AB1.
The following filter circuit W6A branches off the second partial frequency band between 3.3 and 4.3 GHz in its lower section and produces an output voltage U6A by way of a coupler K6A. The signal thus obtained is fed to a modulator M6, 8 which has an heterodyning frequency selected at f06, 8=3 GHz. Thus, also, frequencies of 0.3 through 1.3 GHz are obtained as difference frequencies, i.e. in the second intermediate frequency band.
The third partial frequency band is coupled out between 2.3 and 3.3 GHz by means of the filter circuit W7A, whereby an output voltage U7A is obtained by way of a coupler K7A. The conversion into the second intermediate frequency band 0.3 through 1.3 GHz is undertaken by a mixer M7 whose heterodyning frequency is selected at f07=2 GHz. The resulting signal is then fed to the channel AB1.
An output voltage U8A is obtained from the fourth partial frequency band between 1.3 and 2.3 GHz by way of a coupler K8A. These signals reach a mixer M8 which has an heterodyning frequency f08=2 GHz. Only the sum frequency is permitted to pass through the band pass filter BP8 so that frequencies between 3.3 and 4.3 GHz are fed to the mixer M6, 8 mentioned above. By means of the additional conversion, which is undertaken in the mixer M6, 8, again values between 0.3 and 1.3 GHz are formed in the second intermediate frequency band and are fed to a channel AB2. Thus, the resulting heterodyning frequency amounts to 1 GHz for the fourth partial frequency band.
Therefore, the circuit proceeds in an analogous manner to the circuit illustrated in FIG. 1 such that the partial frequency bands (respectively 1 MHz wide) of the filter group IIA are alternately fed to the channel AB1 (uneven-numbered partial frequency bands) and to the channel AB2 (even-numbered partial frequency bands). The frequency ranges of 4.3-5.3 GHz and 2.3-3.3 GHz from the channel A of the first intermediate frequency band are contained in the channel AB1. However, the frequency ranges of 3.3-4.3 GHz and of 1.3-2.3 GHz reach the channel AB2.
A filter group IIB, illustrated in the lower part of FIG. 2, is constructed in the same manner as the filter group IIA. The individual filter circuits W5B, W6B and W7B have the same frequency ranges as in the filter group IIA, as is illustrated from the frequency values provided in the respective filter block symbols of the drawing. The connecting to the two channels AB1 and AB2 results in the same manner as in the filter group IIA. This means that the uneven-numbered partial frequency bands of between 4.3 and 5.3 GHz and between 2.3 and 3.3 GHz coming from the channel B are contained in the channel AB1, whereas the even numbered partial frequency bands of between 3.3 and 4.3 GHz and between 1.3 and 2.3 GHz coming from the channel B are fed to the channel AB2.
The mixers M5, M7, M8, M6, 8, already mentioned in the filter group IIA, are also utilized for conversion, which is readily possible due to the same partial frequency bands in the filter groups IIA and IIB, and the same second intermediate frequency band of 0.3 through 1.3 GHz, and provides the same heterodyning frequencies. The respective output voltages are referenced U5B-U8B for the filter group IIB.
A circuit constructed in accordance with FIG. 3 is connected to the output terminals 3a, 3b of FIG. 2 and contains two filter groups IIIA and IIIB. The upper filter group IIIA has a chain-connected arrangement of filter circuits W9A-W12A, whereby the partial frequency bands of only 0.2 GHz are coupled out. The distribution of the new partial frequency bands to the filter circuits is selected in the same manner as previously discussed with respect to FIGS. 1 and 2. More specifically, the filter circuit W9A couples out the partial frequency band between 1.1 and 1.3 GHz. The coupled-out frequency band lies between 0.9 and 1.1 GHz in the filter circuit W10A, between 0.7 and 0.9 GHz in the filter circuit W11A, and between 0.5 and 0.7 GHz in the filter circuit W12A, and the last partial frequency band comprises frequencies between 0.3 and 0.5 GHz. The output voltages occurring at the outputs of the filter circuits are branched off by corresponding couplers K9A-K13A and are referenced U9A-U13A, respectively. The received signals reach a plurality of mixers M9A-M12A, whose heterodyning frequencies f09-f012 are respectively selected to have a difference of 0.2 GHz. First of all, there is an upward modulation with the frequency f013A=0.4 GHz in the mixer M13A, and thus the sum frequency (after filtering by a band pass filter BP13A) is converted into the differential frequency in the mixer M11A, this frequency being in the range of 0.1 and 0.3 GHz. Therefore, the resulting heterodyning frequency for the fifth partial frequency band is at 0.2 GHz. A third intermediate frequency band lying between 0.1 to 0.3 GHz occurs at the outputs of these mixers. The sequence of the distribution of the individual partial frequency bands of 0.2 GHz band width each results such that the uneven-numbered partial frequency bands (first partial frequency band of 0.1-0.3 GHz; the third frequency band of 0.7-0.9 GHz; and the fifth partial frequency band of 0.3-0.5 GHz) in sequence are fed to a channel AB11. The even numbered partial frequency bands (second partial frequency band of 0.9-1.1 GHz; and the fourth partial frequency band of 0.5-0.7 GHz) in the sequence are, however, fed to a channel AB21. Therefore, the receiving frequencies lies only in the third intermediate frequency band, i.e. between 0.1 and 0.3 GHz at the outputs 4a and 4b of the two channels AB11 and AB21, respectively.
In the lower portion of FIG. 3 an additional filter group IIIB is illustrated. The structure of the filter group IIIB, which has filter circuits W9B-W12B, is the same as that of the filter group IIIA, and also the respectively processed frequencies (the partial frequency bands and residual frequencies and third intermediate frequency band) result in the same values as in the filter group IIIA. Accordingly, the respective frequencies for the heterodyning frequencies in the modulators M9B-M13B are also completely equal to the respectively analogous referenced mixers M9A-M13A of the filter group IIIA. The distribution of the respective output voltages in the individual partial frequency bands is determined by a plurality of couplers K9B-K13B, in the same manner as mentioned above, to supply the respective voltages U9B-U13B.
A difference in relation to the filter group IIIA exists in the filter group IIIB in that the uneven numbered partial frequency bands (the first partial frequency band of 1.1-1.3 GHz; the third partial frequency band of 0.7-0.9 GHz; and the fifth partial frequency band of 0.3-0.5 GHz) are fed to the channel AB21, whereas the even-numbered partial frequency bands (the second partial frequency band of 0.9-1.1 GHz; and the fourth partial frequency band of 0.5-0.7 GHz) are fed to the channel AB11.
FIG. 4 illustrates circuits for the additional evaluation of the signals of the channels AB11 and AB21 supplied by the circuit of FIG. 3. The channel AB11 having a bandwidth of 100 through 300 MHz is fed in at the terminal 4a. The analysis of this frequency range of 200 MHz bandwidth results in a filter bank A comprising 43 parallel-connected band pass filters BP1-BP43. Each of these band pass filters has a pass band of 10 MHz, the respective center frequency being provided in the corresponding symbolic filter block. The band pass filter BP1 therefore has a pass band of 90-100 MHz. The band pass filter 43, provided for the highest frequencies, has a pass band of 300-310 MHz. The output voltages for the respective filters, after rectification, are referenced UF1-UF43. The fact that the band pass filter BP1 and the band pass filter BP43 actually extend outside of the third intermediate frequency band is based on the fact that even in the border or limiting frequency ranges signal portions can occur by means of the modulation of the received signals.
The channel AB21 is connected to a filter bank B by way of the terminal 4b. The filter bank B is constructed the same as the filter bank A. Therefore, 43 band pass filters are also provided, which filters are referenced BP44-BP86, each of these filters again having a pass band of 10 MHz. The rectified output voltages of the filters are referenced UF44-UF86 in the filter bank B.
The object now is to determine, by means of a corresponding logical linkage of the output voltages U1-U13B obtained from the individual partial frequency bands in the respective filter groups and with the aid of the output voltages UF1-UF86 of the filter banks A and B of FIG. 4, in which frequency band a received signal was originally located. An additional objective is to unequivocally determine the frequency position of signals whose frequency falls into one of the gray zones of the filter groups. The evaluation circuits of FIG. 5 and FIG. 6, which are to adjoin each other with FIG. 6 to the right of FIG. 5, are provided for this purpose.
The different output voltages U1-U13B of FIGS. 1-3 are fed to memories, in particular read only memories, hereinafter referred to, at least partially, by the designation ROM. The respective output voltages either have the value of "0" (i.e. no output signal is present) or the value of "1" (i.e. output signal is present). Each logical "1" reads the appertaining frequency number in a read only memory ROM, i.e. reads a combination of bits which contains the frequency value of the respective resulting overlapping frequency in coded form. Thus, that frequency is determined in the resulting heterodyning frequency which corresponds with the difference of the respective frequency band minus the appertaining common intermediate frequency band.
The respective filter groups of FIGS. 1-3 are referenced by the corresponding Roman numerals and supplementarily by the capital letters in the read only memories ROM. In the upper portion of FIG. 5, the output voltages U1, U3 and U4T (the uneven numbered partial frequency bands from FIG. 1) are fed to a read only memory ROM-I-U, where the letter U respectively stands for the uneven numbered partial frequency bands. The even numbered output voltages U5A and U7A of the filter group IIA, according to FIG. 2, are fed to a read only memory ROM-IIA-U. The even-numbered output voltages U6A and U8A of the filter group IIA, according to FIG. 2, are input to a read only memory ROM-IIA-G, where the letter G refers to the even-numbered partial frequency bands. The uneven-numbered partial frequency bands corresponding to the output voltages U9A, U11A and U13A of the filter group IIIA of FIG. 3 are fed to a read only memory ROM-IIIA-U, whereas a read only memory ROM-IIIB-G receives the output voltages U10B and U12B of the even-numbered partial frequency bands of the filter group IIIB in accordance with FIG. 1.
The read-only memory ROM-I-G illustrated in the lower portion of FIG. 5 receives, in an analogous manner, the even-numbered output voltages U2 and U4N of the filter group I, a read only memory ROM-IIB-U receives the uneven-numbered output voltages U5B and U7B of the filter group IIB, a read only memory ROM-IIB-G receives the even-numbered output voltages U6B and U8B of the filter group IIB, a read only memory ROM-IIIA-U receives the uneven-numbered output voltages U9A, U11A and U13A of the filter group IIIA, and, finally, a read only memory ROM-IIIB-G receives the even-numbered output voltages U10B and U12B of the filter group IIIB.
The frequency number of the read only memory ROM-I-U is conveyed to two parallel adding stages AD11 and AD12. An additional adding stage AD13 is connected to the output of the adding stage AD11, which also receives the frequency number of the read only memory ROM-IIA-U. The adding stage AD12 is followed by an adding stage AD14 to which the frequency number of the read only memory ROM-IIB-G is fed. The output of the adding stage AD13 is connected, in multiple, with a pair of summation stages S11 and S143 (described in greater detail below), and to which the respectively resulting frequency numbers from the filter groups are fed. The result at the output of the adding stage AD14 is connected, in multiple, to a pair of summation stages S21 and S243.
The frequency number of the read only memory ROM-I-G is fed to a pair of parallel adding stages AD21 and AD22, whereby the adding stage AD21 obtains the frequency number of the read only memory ROM-IIB-U at one of its inputs, and the adding stage AD22 receives the frequency number of the read only memory ROM-IIB-G. The output of the adding stage AD21 is fed to an adding stage AD23 which also obtains the frequency number of the read only memory ROM-IIIA-U. The value thus obtained is fed, in multiple, to a pair of summation stages S31 and S343. The output signal of the adding stage AD22 is applied to an adding stage AD24 which also receives the frequency number of the read only memory ROM-IIIB-G. The frequency number obtained after the addition in the adding stage AD24 is fed, in multiple, to a pair of summation stages S41 and S443.
In the right-hand portion of FIG. 5, the logical linkage of the information obtained from the read only memories is illustrated with the voltage values UF1-UF43 obtained from the band pass filters BP1-BP43 of the filter bank A, whereby--in order to simplify--only the processing of the first output signal UF1 and the last output signal UF43 is specifically referenced.
The absolute magnitude of the voltage value is determined by each voltage value UF1-UF43 and converted into a digital word, a plurality of analog/digital converters DPM1-DMP43 being provided for that purpose. This level information is fed to evaluator RE, in which the evaluation of the individual information is carried out.
The individual voltage values UF1-UF43, moreover, are respectively fed to a read only memory ROM1-ROM43 as 1-bit information items (voltage present="1", voltage missing="0").
A logical "1" calls the frequency number, characterizing the respective frequency band, in the appertaining read only memory ROM1-ROM43, this frequency number then being made available at the output. A line of summation devices is connected to the output of each of the read only memories ROM1-ROM43, these summation devices respectively comprising four summation stages S11, S21, S31 and S41 (for the read only memory ROM1) and S143, S243, S343, S443 (for the read only memory ROM43), these stages being connected in multiple to the output of the appertaining read only memory.
The summation stages S11-S443 form an overall frequency number from the two respectively adjacent frequency numbers, this overall frequency number being fed to the computer RE. This overall frequency number indicates the frequency value of the respectively received signal. In order to obtain unequivocal results, the adding stages AD11-AD24, and the summation stages S11-S443 must be constructed such that they only provide sum output signals when the two inputs are simultaneously engaged, or are engaged within a specific time (dependent upon processing time). Thus, only that line of the summation matrix, corresponding with the respective signal paths, receives a frequency number because only in such a line are three filter groups sequentially engaged. As only one column (only one receiving signal respectively presumed) is engaged by one frequency number, the summation stage Snm which is at the intersection of an engaged line and an engaged column precisely indicates the correct frequency value of the received signal and supplies the overall frequency number to the evaluator RE.
FIG. 6 illustrates the logical linkages of the information obtained by the filter bank B with the various output voltages of the individual filter groups. The construction is completely analogous to the embodiment illustrated in FIG. 5. Thus, for the output voltages UF44-UF86 the respective analog/digital converters DPM44-DPM86 are provided which transmit the information to the computer RE by way of the signal amplitudes.
Moreover, by each engaged band pass filter BP43-BP86 the appertaining frequency number is formed in the read only memories ROM44-ROM86 and is transmitted to the summation stages S144-S486. Respectively, four such summation stages are connected in a multiple and arranged as a column.
The read only memories ROM-I-U, ROM-IIA-U, ROM-IIA-G and the read only memories ROM-I-G, ROM-IIB-U and ROM-IIB-G receive the same frequency information as the correspondingly referenced components in FIG. 5. They are connected with the adding stages AD11, AD12, AD21 and AD22 in the same manner as in FIG. 5.
For the remaining read only memories, however, the following distribution of the output voltages result:
ROM-IIIA-G--output voltages U12A, U10A
ROM-IIIB-U--output voltages U13B, U11B, U9B
ROM-IIIA-G--output voltages U12A, U10A
ROM-IIIB-U--output voltages U13B, U11B, U9B.
The read only memory ROM-IIIA-G is connected to the adding stage AD13 which supplies the resulting frequency number to the summation stages S144 and S186 of the summation matrix. The frequency numbers reach the adding stage AD14 from the read only memory ROM-IIIB-U, and supplies the result to the second line of the summation matrix, that is to the summation stages S244 and S286. The frequency numbers of the read only memory ROM--IIIA-G are applied to an adding stage AD23 from which the resulting frequency number is transmitted to the summation stages S344 and S386, the third line of the summation matrix. The frequency numbers of the read only memory ROM-IIIB-U are applied to the adding stage AD24 which, in turn, supplies the summation stages S444 and S486 of the summation matrix.
In the summation matrix provided for the filter bank B, also, only one single column (with only one received signal) is engaged by one of the read only memories ROM44-ROM86. In the same manner, one resulting frequency number is present in only one single line, this frequency number originating from, respectively, three successive filter groups. Thus, there also, only one single summation stage is actuated twice, and results in one overall frequency number characterizing the frequency of the received signal.
For the explanation of the operating characteristic of the circuit of the present invention, several examples of receiving frequencies are subsequently provided and the appertaining results are illustrated in tabular form.
TABLE I ______________________________________ Unique Reception (Outside of Gray Range) Receiving Frequency: 8.4 GHz Volt- Resulting IF Chan- Filter age Memory Freq.f0r (GHz) (GHz) nel ______________________________________ W3 U3 ROM-I-U 7 1.4 A W7A U8A ROM-IIA-G 1 0.4 AB2 W12B U13B ROM-IIIB-U 0.2 0.2 AB21 ______________________________________
From the sum of the f0r, the value of 8.2 GHz is the result, in toto. To this value, the frequency value of 0.2 GHz is added from the filter band B in the matrix, in accordance with FIG. 6, so that, in toto 8.2+0.2=8.4 GHz which is the resulting received frequency. The second line of the summation matrix, in accordance with FIG. 6, is activated because at the output of the adding stage AD14, a frequency number (8.2 GHz) occurs which leads to a coincidence in the column corresponding with the band pass filter of 200 MHz, and thus is also engaged, and supplies an overall frequency number of 8.4 GHz to the evaluator RE. In FIG. 5, however, as can easily be determined with the aid of the engagement of the read only memory, no engagement of a line takes place, and thus no indication is provided in the evaluator RE. No column is engaged, in FIG. 5, also because the filter bank A is not activated (i.e. no signal in the channel AB11).
TABLE II ______________________________________ Reception In The Gray Range Receiving Frequency: 12.3 GHz Volt- Resulting IF Chan- Filter age Memory Freq.f0r (GHz) (GHz) nel ______________________________________ (a) Path 1 W2 U2 ROM-I-G 11 1.3 B W7B U8B ROM-IIB-G 1 0.3 AB2 W12B U13B ROM-IIIB-U 0.2 0.1 AB21 (b) Path 2 W3 U3 ROM-I-U 7 5.3 A W5A U5A ROM-IIA-U 4 1.3 AB1 W9A U9A ROM-IIIA-U 1 0.3 AB11 ______________________________________
For the path 1 in FIG. 6 results an engagement of the fourth line by means of the frequency number of 12.2 GHz at the output of the adding stage AD24. Together with the band pass filter from the filter bank B for 100 MHz center frequency (engagement of one column) results in their overall frequency number of 12.3 GHz in the evaluator RE. By means of the filter bank A (engagement of the band pass filter for 300 MHz) and the engagement of the first line of the summation matrix in accordance with FIG. 5, there also the overall frequency number of 12.3 GHz is formed in the evaluator RE.
It should be pointed out that the number of the successive filter groups primarily depends on the width of the frequency band to be analyzed, and how fine the desired final resolution. The filter circuits illustrated in FIGS. 1-3 can therefore be varied in a multiple manner. In the extreme case, (in accordance with the filter group I of FIG. 1) the filter banks A and B of FIG. 4 can be directly connected to the two transmission channels A and B (however, with a correspondingly altered operational frequency range, namely of 1.3-5.3 GHz). On the other hand, it is also possible to provide additional filter circuits. It is only to be observed that the alternating distribution of the output signals to the transmission channels, and the overlapping-free transmission paths are to be maintained.
Although we have described our invention by reference to particular illustrative embodiments thereof, many changes and modifications of the invention may become apparent to those skilled in the art without departing from the spirit and scope of the invention. We therefore intend to include within the patent warranted hereon all such changes and modifications as may reasonably and properly be included within the scope of our contribution to the art.
1. In a circuit arrangement for analyzing and determining the frequencies of signals received in a wide frequency band, of the type wherein a filter bank is cooperatively connected with an evaluation circuit, the filter bank has a plurality of band pass filters with adjacent pass bands which cover an evaluation frequency range which is very small in relation to the overall frequency band, wherein the received signals are processed by means of a filter circuit having a plurality of adjacent pass bands and a plurality of modulators connected thereto to convert the signals into the evaluation frequency range of the filter bank, and wherein the individual heterodyning frequency values are added in the frequency analysis, the improvement therein comprising:
- a pair of transmission channels connected to said modulators such that modulators associated with first alternate frequency bands are connected to one of said channels and the modulators associated with second alternate frequency bands are connected to the other of said channels;
- a pair of filter banks connected to respective ones of said transmission channels; and
- means connected to said filter circuit and to said filter banks and responsive to signals in the respective pass bands to provide respective output signals characterizing the signals in those pass bands, whereby each received signal traverses only one overlapping-free transmission path if it falls in only one pass band and traverses two overlapping-free transmission paths if it falls into two adjoining pass bands to result in unequivocal frequency values indicated by said output signals for evaluation.
2. The improved circuit arrangement of claim 1, wherein said filter circuit comprises a first filter group, and a plurality of pairs of second filter groups, each of said filter groups including an input connected to only one transmission channel, respectively, and an output connected, alternatingly, to two transmission channels, said filter groups thus being connected in a chain with the last filter groups connected to said two filter banks.
3. The improved circuit arrangement of claim 2, comprising a plurality of read only memories each connected to and actuated by selected combinations of said output signals, a read only memory which contains the appertaining resulting heterodyning frequency being activated to provide a frequency indicating number for evaluation, and a plurality of adding circuits connected to said read only memories for adding the frequency numbers emitted by said read only memories.
4. The improved circuit arrangement of claim 3, comprising a summation matrix connected to said adding circuits and including a plurality of summing circuits arranged in columns and rows with the summing circuits of an individual row connected to the same adding circuit, a plurality of additional read only memories each having an output connected, in multiple, to each summing circuit of a column, and having an input connected to receive a respective output signal from said filter banks and responsive thereto to indicate the presence of a frequency being analyzed in the respective pass band, whereby the appertaining frequency number is produced when a signal is engaging the filter bank in the respective pass band, and the summing circuits activated by said adding circuits and said additional read only memories provide the frequency numbers which result from substantially coincident engagement of columns and rows of the matrix.
5. The improved circuit arrangement according to claim 1, wherein said filter circuit and said filter banks comprise individual filters whose adjacent pass bands overlap one another with respect to pass attenuation, in such a manner that received signals which lie in the gray range and fall into two pass bands are attenuated as little as possible in at least one of the appertaining pass bands.
- A. Galvin, A Sequential Detection System for the Processing of Radar Returns, Proceedings of IRE, vol. 49, 1961, No. 9, pp. 1417-1423.
Filed: Nov 17, 1977
Date of Patent: Jul 12, 1988
Assignee: Siemens Aktiengesellschaft (Berlin & Munich)
Inventors: August Weber (Unterpfaffenhofen), Erich Mayerhofer (Taufkirchen), Klaus Weis (Neuried), Guenter Beyer (Geisenbrunn)
Primary Examiner: Theodore M. Blum
Law Firm: Hill, Van Santen, Steadman & Simpson
Application Number: 5/854,403
International Classification: G01R 2316; G01R 2702;


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Q40:high performance 500A AC current clamp CT sensor
The Q40 Series AC Current clamp ct sensor are high performance models, and use a custom nickel core to improve accuracy and phase shift on the low end of the measurement range.
UL, CE, CNAS mark;
Installation: clamp type design;
Holding wire diameter: φ40mm;
Measurement range of 1mA to 600A AC;
Lower phase shift for power measurement;
Improved ergonomic design and easy operation;
Designed to EN 61010, 600V CAT III safety standard;
Suitable for 10A, 50A, 100A, 500A cable online measurement;
Basic parameters :
Ratio:1000: 1 or 2000:1
Accuracy class:0.1%, 0.2%, 0.5%
Primary current:0 - 500A AC
Secondary current:0 - 500mA AC or 0-5V AC (customized)
Max. Cont. Input current:600A
Added (m.d.y): 06.23.2022
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The combination of visual and inertial sensors has proved to be very popular in MAV navigation due the flexibility in weight, power consumption and low cost it offers. At the same time, coping with the big latency between inertial and visual measurements and processing images in real-time impose great research challenges. Most modern MAV navigation systems avoid to explicitly tackle this by employing a ground station for off-board processing. We propose a navigation algorithm for MAVs equipped with a single camera and an IMU which is able to run onboard and in real-time. The main focus is on the proposed speed-estimation module which converts the camera into a metric body-speed sensor using IMU data within an EKF framework. We show how this module can be used for full self-calibration of the sensor suite in real-time. The module is then used both during initialization and as a fall-back solution at tracking failures of a keyframe-based VSLAM module. The latter is based on an existing high-performance algorithm, extended such that it achieves scalable 6DoF pose estimation at constant complexity. Fast onboard speed control is ensured by sole reliance on the optical flow of at least two features in two consecutive camera frames and the corresponding IMU readings. Our nonlinear observability analysis and our real experiments demonstrate that this approach can be used to control a MAV in speed, while we also show results of operation at 40 Hz on an onboard Atom computer 1.6 GHz.
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A circular electric current, perpendicularly bisected by a field plane, was modelled as a continuum of pairs of distinct Cartesian component elemental magnetic vector charges normal and parallel to the field plane. The Cartesian elemental magnetic vector charges normal to the field plane pair up into Cartesian elemental magnetic dipoles with intra-dipolar displacements parallel to the plane. These dipoles generate the overall magnetic vector potential at the field point. The Cartesian elemental magnetic vector charge components parallel to the field plane form Cartesian elemental magnetic dipoles (with intra-dipolar displacements perpendicular to the field plane) which individually and collectively contribute nothing to the overall magnetic vector potential. Each continuum of these two sets of Cartesian elemental magnetic dipoles independently yields the traditionally renowned “magnetic dipolar moment of a circular current”. However, together their two distinct magnetic fields, as well as their two distinct magnetic torques, constitute the circular current’s overall magnetic field and the total magnetic torque respectively. These results reconcile only if the magnetic dipolar moments of both sets are endorsed, that is a circular current of any spatial size is a continuum of pairs of distinct Cartesian elemental magnetic dipoles and that consequently its overall magnetic dipolar moment is numerically twice the traditional value. In addition the customary ad hoc definition of the magnetic dipole moment of a current loop is deceptively erroneous, thus prompting a review of many relations involving it. These include the magnetic torque and magnetic field generated by it, and the classical magneto-mechanical ratio.
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