current sense transformer in cascade converter - Page 1
Forum section: electronics / projects

author: phenol
author_rank: Regular Contributor
timestamp: January 14, 2016, 08:59:59 pm
content: Im working on a 12V-440VDC converter. The topology of choice is current-fed push-pull with buck stage preregulation based on LM5041 and a handfull of other auxiliary ic's. It employs peak current mode control with cycle-by-cycle current limit should things spin out of control.As such, the current flowing into the high side buck stage mosfet is monitored via a current sense transformer in a unipolar fashion, which basically means that the available reset time diminishes more and more as duty cycle increases, such as during load transients or low batt voltage, or both, and can become as short as 250-300ns. Switching period of the buck stage is 15us.The current sense transformer /cst/ signal starts to droop and assume negative slope as CST core saturation ensues. It seems unable to reset itself resonantly withing the available 250ns. The rectifier diode on the CST output is BYV26c and reset voltage (reverse voltage for the diode) peaks to -400V - -500V under extreme loading (which further poses isolation issues and such), but the core nevertheless saturates within couple of switching cycles and the sensed current waveform becomes distorted. In turn, this destabilizes the inner current control loop of the controller.Replacing the fast recovery byv26 with a no recovery schottky 600v SiC diode vastly improved the situation and the CST now resets itself comfortably within the short time available to do so with -200 or so volts across the SiC part and no drooping is visible.The lack of reverse recovery current in SiC diodes to counteract resonant CST reset did help a lot, but i would appreciate suggestions on alternate CST reset methods when only unipolar sensing is possible.In a previous conventional pushpull design i just ran the two legs of the transformer (instead of the center tap) thru one common cst in antiphase which made it bipolar and capable of very high duty cycles without saturation, but the topology now is somewhat different.....

author: Richard Head
author_rank: Frequent Contributor
timestamp: January 15, 2016, 07:10:21 am
content: I would be inclined to consider a resistor in the negative lead and a high speed op-amp. If it's a low power application an op-amp may not even be required.I know it's not as efficient but it's cheaper and cleaner than using a CT.But, why is a two stage converter required?Below 75W I would go for a simple flyback converter. I know the peak currents get huge so it depends on the power level.

author: T3sl4co1l
author_rank: Super Contributor
timestamp: January 15, 2016, 07:25:26 am
content: Circuit?DCM sounds like the exact opposite of what you'd want for a current fed converter...Tim

author: Richard Head
author_rank: Frequent Contributor
timestamp: January 15, 2016, 08:04:36 am
content: Current fed works well when dealing with high voltage outputs that have output diodes with poor reverse recovery time. DCM flyback is ideal for low power low cost applications. Above about 500-1000VDC inter-winding capacitance seems to become a problem, I've found.

author: phenol
author_rank: Regular Contributor
timestamp: January 15, 2016, 09:20:15 am
content: The circuit as a broad approximation is similar to what they have in the datasheet of LM5041, with different power transformer design, of course. The output power is 500W. I've found the exact same problem with conventional pushpull as the one Richard mentioned--xfrm interwinding capacitance plus secondary side ringing cause extremely wide leading edge spike in the primary current, requiring special leading edge blanking efforts, and what is worse--secondary side rectifier ringing in tandem with the output power inductor (~2mH). Unless snubbed, the bridge rectifier diodes experience ringing in excess of 1kV....even with SiC diodes. Passive snubbing burns a lot of power.With the cascaded topology, regulation is done in the buck stage and the pushpull part is simply an 'ideal' dc transformer. Thus, i can use lower voltage mosfets and 600V rated Si rectifiers.I cannot use low side sensing, the current has to be monitored on the buck switch, not the sync switch. I did actually try a 0.5milliohm resistor in the buck switch drain and an array of high speed op amps and exotic high side current sensing ic's, but the strong common mode noise and switching transients/fields located in this area (plus the less than optimal layout of current sensing chips during trials) rendered the output signal useless. I even tried a rogowski coil-based sensor--Si8519--but then again the output was somewhat jittery. ANything with active signal conditioning inside seems to suffer in performance in the hostile electromagnetic environment around it.The two screenshots show the CST output signal as taken directly across the burden resistor with 1. BYV26C diode and 2. C3D1P7060Q SiC schottky under equivalent loading. As you can see, in 1. as soon as the duty cycle becomes large enough (Batt voltage going lower), CST fails to reset and saturates. This does not happen with the sic diode. The peak current is slightly less than 60A.The actual current waveform going into the PWM controller is different because there is slope compensation ramp added to it.

author: Richard Head
author_rank: Frequent Contributor
timestamp: January 15, 2016, 10:23:41 am
content: PhenolIt sounds like you may have serious layout/grounding issues that are making you tear your hair out. I know the feeling. Your star point should be the bottom of the freewheel diode.I firmly believe that a single stage push-pull PWM converter should be able to do the job. Also, I wouldn't be too ambitious about the operating frequency. Remember that switching losses are proportional to frequency so being conservative here pays dividends.How have you addressed your transformer leakage inductance? Have you interleaved primary and secondary? It makes a huge difference to the leakage. With the large turns ratio required the coupling will not be great unless interleaved.For a transformer design I normally aim for a peak flux density of about 100mT when working at 100Khz for ferrite transformers. This results in lowest total losses (copper + core). Also, don't use an unnecessarily large transformer as this introduces unnecessary leakage which will then have to be snubbed.

author: phenol
author_rank: Regular Contributor
timestamp: January 15, 2016, 10:49:19 am
content: Hi Richardthe actual pushpull stage switching frequency is half of that of the buck stage. I can go lower, but doing so leaves the CST more time to saturate...as you can see, my woes revolve around that one component.I have a conventional single stage pushpull converter with LM5030 of similar output power and voltage, but it suffers from what i described in my previous post. The output snubber dissipates a lot of power and occupies a lot of space, efficiency is altogether lower that this one, and i just have to use 1200V diodes on the output. It works fine, but....meh.Grounding is as you say-commoned at the sync source point. I have a small signal ground where the pwm controller is and it's all fine and dandy. It is also somewhat away from power switches and magnetics, which was not the case with the high side current sensors i played with. Because of the small sense resistor, they had to have gain of at least 10. Plus, AC CMRR would be quite poor as far as fast switching transients are concerned. The result--what looks like primary current waveform with superimposed spikes amplified many times more that the difference signal. I'll just have to go with a cst at this stage.As for the power transformer, winding stackup is as follows: 1 layer secondary-1/2primary-center tap-2/2 primary- 1 layer secondary. The primary is copper foil. With the secondary shorted, the measured primary inductance is less than 100nH.Anyway, i have added active recovery clamp there. Basically a small buck converter that turns on and feeds energy back to the input when the peak voltage across the pushpull switches exceeds a certain voltage due to leakage spiking. This energy can be used to power a fan, for instance

author: Richard Head
author_rank: Frequent Contributor
timestamp: January 15, 2016, 11:48:34 am
content: PhenolYou are trying to achieve the impossible with the CT and the buck at very high duty cycles. Is the CT in the drain or source lead? Drain lead is better as dv/dt is less and gate charging current is absent.You could also feed the freewheel diode lead through the CT (in opposite direction)and place a full wave bridge rect. on the CT output. The CT waveform will then include the freewheel diode current but more importantly it will force the core flux to remain balanced. This may give to a little more flux headroom and allow you to use lower voltage diodes. Also, ensure you CT core isn't too small as that will also cause droop. Try to keep flux density in CT below 50mT.Anyway, I still believe you'll get better results from a single stage topology. Best of luck.

author: phenol
author_rank: Regular Contributor
timestamp: January 15, 2016, 12:37:33 pm
content: thanks for the suggestion, combining the two currents thru the same ct may work.The actual max duty cycle is limited by the pwm controller with minimum off time ~250ns, so my objective was to get the ct to reset within that time. I may be able to also stretch that offtime somewhat and provide more reset time.The only reason why i went with this cascaded topology was secondary side rectifier overshoot and the associated difficulty snubbing it. There is no ringing now, completely flat tops. I can use 40V or even 30V rated devices with sub-milliohm channel resistance and much lower gate charge...

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 15, 2016, 01:12:17 pm
content: I'm not seeing the advantage of cascading a buck preregulator here, either; mainly because any voltage drop in the buck switch(es) will have an outsized effect on efficiency.I would stick with a conventional peak current-mode push-pull if the input supply doesn't vary too much (e.g. - 11-13V) and the load isn't too abusive, otherwise, a Weinberg (flyback current-fed push-pull) will give you extreme robustness, with no output inductor and the ability to steer ripple current away from the input or output at the expense of needing a somewhat-tricky-to-optimize flyback transformer. See, for example, the TI demo board at:http://www.ti.com/tool/PMP4627If you really want to stick with a buck current-fed push-pull for whatever reason, then be assured that you can place the primary windings of the current transformer in series with the two primaries of the push-pull (ie - one turn for each PP primary wound in anti-phase), you just won't pick up buck choke charging current (ie - when the PP switches are both on). That is to say, in this location the CT will only pick up the load current, but as long as the buck choke doesn't saturate and the peak-to-peak ripple current doesn't come close to the buck switch (or FWD!) rating then this is perfectly acceptable.Otherwise you can consider a zero current transition interleaved buck stage as described by Illic, Maksimovic in a 2007 IEEE paper (see DOI "10.1109/TIA.2007.908175"); in this type of interleaved buck the duty cycle of each switch is summed together to get the output, and thus each switch sees the same peak and average current (as compared to conventional interleaving, as commonly used in modern CPU core voltage regulators, for example).Or you can eat the higher parts cost of a Hall-effect (or GMR, etc.) type current sensor. Allegro makes some nice parts with an integrated bus bar good for up to 50A or so.Etc. and so on... lots of ways to skin this cat.

author: phenol
author_rank: Regular Contributor
timestamp: January 15, 2016, 02:26:14 pm
content: Criss-crossing the PP xfrm leads thru the CT is what i did in the conventional push-pull version. It looks awkward and it's hard to do whilst keeping leads short, but it works a treat.As for Allegro's and other hall sensors, they are just too slow to react, particularly in transient overload conditions or shorts, for adequate pulse by pulse current limiting. I may also consider some soft switching topologies like LLC, for instance, in the distant future...for now this would have to do.

author: Richard Head
author_rank: Frequent Contributor
timestamp: January 15, 2016, 05:37:39 pm
content: I think, even if it's only for academic value, I would try to find out why there is so much ringing across the output rectifiers of the single stage push-pull unit. Did you bypass the primary supply adequately right at the transformer center tap using low ESR caps? are you switching on the MOSFETS too fast for the diodes?Also, with current mode control if you choose a low ripple current (10-15%)in the output choke it is very difficult to extract the small ramp, particularly in the presence of noise. Generally the best choice is around 30% I've found.Incidentally, are you using a short wire measurement (scope probe spring grounded to probe tip ground) technique?

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 15, 2016, 05:49:28 pm
content: Quote from: phenol on January 15, 2016, 02:26:14 pmOne of the primary advantages of a current-fed topology is intrinsic robustness against output shorts because the inductor limits how quickly current can ramp up, so you don't really need fast current limiting... The conventional push-pull does need fast, pulse-by-pulse current limiting, but not the current fed version.

author: phenol
author_rank: Regular Contributor
timestamp: January 15, 2016, 07:09:57 pm
content: yes, indeed, it limits the rate of rise, but if there is too much delay, the actual pulse width during continuous short would become rather too long before it finally trips the first cycle termination comparator. This essentially results in switches meltdown. i slowed it down on purpose once and the buck switch desoldered itself and fell on the floorthere is a second comparator, which when tripped, results in hiccup operation, but the short cct current never rises quickly enough to reach that second threshold.Quote from: MagicSmoker on January 15, 2016, 05:49:28 pm

author: phenol
author_rank: Regular Contributor
timestamp: January 15, 2016, 07:33:24 pm
content: Quote from: Richard Head on January 15, 2016, 05:37:39 pmyes, im using a spring on the probe tip ground. it's a 600MHz scope and it picks all sorts of crap with long gnd leads, unless bandwidth limited to 20MHz.the gnd spring itself is now badly deformed due to multiple accidental shorts.as for secondary side diode ringing, it's a resonant effect of diode capacitance during turn off and filter choke inductance+interwinding capacitance. i resorted to using SiC diodes, and it did yield some improvement, but the ringing was still there... and well over 1000V, only shorter.

author: Jay_Diddy_B
author_rank: Super Contributor
timestamp: January 15, 2016, 08:02:29 pm
content: Phenol,I think that you would benefit from changing the turns ratio in the power transformer so that the buck stage does not need to go to 95% duty cycle, then there will be plenty of time to reset the current sense transformer.You could use UVLO to shutdown if the input voltage is too low and requires very high duty cycle.Regards,Jay_Diddy_B

author: phenol
author_rank: Regular Contributor
timestamp: January 15, 2016, 08:20:46 pm
content: I tried that, but the feedback loop blasts full duty cycle during load trsnsients as it attempts to recover the voltage. I guess I'll have to clamp the max duty cycle somehow

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 16, 2016, 01:06:34 pm
content: Quote from: phenol on January 15, 2016, 07:09:57 pmThis doesn't make any sense... Absent other helpful information like a schematic of what you actually built and, ideally, some scope shots of it in action, this doesn't sound like a buck current-fed push-pull; or, perhaps, it could be that the push-pull switches don't have overlapping on times (leading to avalanche failures all over the place). Note that the LM5041 can be configured so that there is a dead time between on periods of the push-pull switches, or an overlapping of the on periods, depending on the connection of a resistor from the TIME pin on the LM5041 to the REF pin (dead time) or GND (overlap).

author: phenol
author_rank: Regular Contributor
timestamp: January 16, 2016, 01:37:51 pm
content: i was talking about an earlier pushpull single stage design of similar output voltage and power. it is the one with severe rectifier ringing, which is the reason why I went with current fed pp, the subject of this thread.LM5041 is configured correctly with slight pp stage overlapping and the rectifier diodes do not have any overshoot on them. as for the meltdown of the buck switch, it was caused by a overly slow current sense filtering and delay... the excessive power had to be dissipated somehow and the mosfet gave up the ghost.the screenshot shows the buck switch current dieectly on cs pin of LM5041 with a shorted output. the inductor is ~1.5uH, the CT output is not filtered. as you can see, it takes the current some time to ramp up and cause the first comparatpr to terminate the pulse and that the rate of rise is just too slow to trip the second comparator and rearm the pwm chip slow start circuit. in my understanding the chip is doing its job

author: T3sl4co1l
author_rank: Super Contributor
timestamp: January 16, 2016, 03:27:32 pm
content: Well the LM5041 datasheet doesn't show current sense transformers, or even shunt resistors, and it shows a "passive" synchronous rectifier. So, I don't really know what you're doing.Which is why I ask first... schematic, please.And layout would be good, too. And transformer windup.Tim

author: phenol
author_rank: Regular Contributor
timestamp: January 16, 2016, 03:48:22 pm
content: T1 on pg14 of the manual. can't provide schematics bc im travelling and I only have my phone with mepp xfrm stackup: 1 layer 1/2 secondary-1/2primary copper foil-center tap-2/2 primary copper foil-1 layer 2/2 secondary. the two secondaries are connected in series. transforation ratio is around 48

author: Jay_Diddy_B
author_rank: Super Contributor
timestamp: January 16, 2016, 04:12:35 pm
content: Hi,Let me help out a little.Here is the full schematic:And the CST section (zoomed):And here is the current transformer used byNationalTI on their EVAL board:Regards,Jay_Diddy_B

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 16, 2016, 11:16:51 pm
content: Quote from: phenol on January 16, 2016, 01:37:51 pmAt no point have I addressed the secondary rectifier ringing, so this is a non-sequitur. That said, rectifiers ring during turn-off because of a resonant tank formed between their junction capacitance and the stray inductance of their wiring to the transformer as well as the leakage inductance of the latter.Quote from: phenol on January 16, 2016, 01:37:51 pmArgh... This is what makes no sense! If you short the output of a buck converter it is not the buck switch that dies from overcurrent, it is the freewheeling diode, and even then that will only happen if the minimum on time of the converter is too long. The minimum on time of the LM5041 is 250ns, so at 12V input with a 1.5uH buck inductor the worst case dI/dt is ~8A/us, which should give the LM5041 plenty of time to act on an overcurrent condition. Furthermore, the application example on p14 of the datasheet shows the current transformer before the buck switch but after the input reservoir capacitors, probably because the current here is pulsating, and thus measurable with a CT, but note that in this location the freewheeling diode is unprotected against persistent output shorts or inductor saturation.But, like I said already, all of these things result in a failure of the freewheeling diode, not the buck switch, so there is something else going on if the latter is, indeed, failing.

author: Richard Head
author_rank: Frequent Contributor
timestamp: January 17, 2016, 10:45:23 am
content: Also, peak current mode control is great for cycle by cycle current limit but not for final current limiting. The reason is that during a short circuit condition on the output the duty cycle pulls right back to minimum and the output inductor is resetting into an almost short circuit. That means that it's dI/dt is very shallow and the short circuit current is now sitting at the peak of the inductor ripple current, not the average current. If you have an inductor ripple current of 30% then your current limit point will be 15% too high. Ideally you require a second current loop after the output caps that pulls back on the output when the current limit is reached. Also, if you are using slope compensation on the current ramp the peak current trip point is duty cycle dependent. That has the effect of causing the current limit point to be much higher at narrow duty cycles than at wide duty cycles. Another reason to have a second current loop.

author: phenol
author_rank: Regular Contributor
timestamp: January 17, 2016, 03:46:17 pm
content: Thanks to everyone for their input.The reason why the buck switch died first may have been the smaller copper area under its drain. The freewheeling 'diode' is another mosfet. Ive added thermal shutdown with a ptc and it stops the pwm controller before things spin out of control, but that's only if the current in the HS switch doesn't exceed 100A. The thermal lag/inertia between the ptc and the switches may cause trouble if current limit is higher.I'm going to try different buck inductors, less and more than 1.5uH, and see what happens when the output is shorted

author: T3sl4co1l
author_rank: Super Contributor
timestamp: January 17, 2016, 04:43:28 pm
content: Quote from: Jay_Diddy_B on January 16, 2016, 04:12:35 pmThanks Jay, but that's for ~50V input, not 12V. And 2.5V output, not 440V.So, still waiting... and layout too, if possible?If the current transformer arrangement is at least representative, then no, you simply can't use a CT for high duty cycles. That much is obvious. If you need DC operation, use a shunt and a fast current sense amplifier.To reduce duty cycle, raise drive current and the transformer ratio. Running on the bleeding edge is bad for input range, anyway.Tim

author: station240
author_rank: Unknown
timestamp: January 19, 2016, 09:01:13 pm
content: I'm wondering if this minor redesign would fix the poor switch off time.Idea is any reverse current through the CT (switch off spikes etc) drain the 470pF capacitor.Some components removed for clarity/drafting reasons. In reality all I did was rearrange existing components.Let me know if you get this PSU working correctly. There is almost no information on designs for "ultra wide range" SMPS, as too many are using Jelly Bean 95-265V parts.I have to design a supply of my own soon, though mine has a lower min/max voltage ratio.

author: phenol
author_rank: Regular Contributor
timestamp: January 20, 2016, 07:57:28 am
content: Well, the negative going spikes is what actually resets the transformer and the lower the interwinding and parasitic capacitances on that node, the better...so, if anything, adding more capacitance there would only make things worse.The CT is hand-wound on a toroid core in a single layer just because of this. Even so, it would not reset itself without a SiC schottky diode (no reverse recovery charge) or a couple of serially connected 100V Si schottky doides.Obviously, CT reset is doomed during excessively large duty cycles, but the internal design of LM5041 inhibits 100% duty cycle, thus leaving around 250ns worst case for CT reset, which, at the current switching frequency and in tandem with a SiC diode, seems to work.