LLC vs LCC Converters for High Output Voltage - Page 1
Forum section: electronics / beginners

author: state_of_flux
author_rank: Contributor
timestamp: May 10, 2019, 02:39:05 pm
content: Hello all,I am currently designing a power supply. The aim is to build a solution that minimises the volume, weight, and size of the solution. I am currently investigating the use of resonant converters, such as series, parallel, hybrid resonant (LLC and LCC) converters, and full bridge phase shifted converters.I understand that the LLC converter leads to the best efficiencies and thus have improved power densities, when compared to LCC. And incorporation of the magnetic components into the transformer also help to reduce size and weight of the solution.My question relates to deciding which could be utilized to its best potential for my application, which is a high output voltage (1-10kV range, 500W+) with dual outputs. I am leaning towards the use of the LLC due to the aforementioned issues, however I often see LCC implemented for high voltage, low current outputs whereas LLC for low voltage, higher current outputs (with synchronous rectification, for example). There are a select few papers in the literature that utilize LLC converters successfully in high output voltage applications - but the scarcity of these kind of makes me err on the side of caution. I have also investigated phase shift converters however their soft switching capabilities seem to be less impressive than LLC and again seem to be used for high current output applications.I know that the winding capacitance of the transformer in a high output voltage design tends to be large - therefore suggesting the possible use of the LCC converter to include this in resonance. However, I am trying to decide whether the issues this capacitance would cause to circuit performance would outweigh that of the benefits of increased power density and efficiency overall.Hope someone can point me in the right direction.Thanks in advance

author: MagicSmoker
author_rank: Super Contributor
timestamp: May 10, 2019, 04:01:56 pm
content: I wouldn't attempt an LLC (or LCC) converter as a total SMPS novice, but I REALLY wouldn't attempt a phase-shifted full bridge, as it deceptively difficult to get working properly, and tends to blow up lots of expensive switches along the way. LLC is probably the most forgiving of the 3 you listed, and is a decent choice for HV outputs on account of it incorporating the leakage inductance into normal operation, but it might not be the best depending on the type of load. Are you looking for more of a constant current type output - which is very common for HV supplies - or do you need precise voltage regulation?

author: state_of_flux
author_rank: Contributor
timestamp: May 10, 2019, 04:57:02 pm
content: Hello MagicSmoker,Thanks for your input! I did see a lot of documentation mentioning how phase shifted design can be troublesome. In this application the regulation of the output voltage is more important. Furthermore it is desirable to operate the converter at a fixed frequency, most likely above resonant frequency to obtain soft commutation of the switches. Hope this information can help in giving more assistance.

author: MagicSmoker
author_rank: Super Contributor
timestamp: May 10, 2019, 06:30:53 pm
content: You can achieve soft switching above or below resonance, so don't let that concern dictate your design. Whether the LLC or LCC topology is the better choice depends more on if stray inductance or capacitance is more of a problem in your transformer design. Using a split core bobbin to separate the primary and secondary is great for isolation, but exacts a terrible price in leakage inductance; conversely, interleaving the primary and secondary drastically cuts leakage but (generally, not always) results in higher stray capacitance - especially between primary and secondary.However, if you need the converter to operate at a fixed frequency then neither LCC or LLC is appropriate, and you might want to look into using auxiliary switch and LC networks to effect lossless transitions. For example, the active-clamp forward is a good choice for this power level and while it mainly benefits switch turn-off (so best for IGBTs) it can be coaxed into achieving lossless (or reduced loss) turn-on, which is best for MOSFETs.

author: state_of_flux
author_rank: Contributor
timestamp: May 11, 2019, 02:43:51 pm
content: That makes sense, the transformer parasitics are what were concerning me most with the resonant designs. Since the output voltage in my case is large, the stray capacitance tends to be more of an issue than the leakage inductance (I believe). I find it difficult to decide whether the leakage inductance or the stray capacitance would be more of an issue without first designing the transformer - but then don't you need to know which topology to use, before you can design the transformer? This is something that does confuse me.I do seem to see some applications in my research area use different control methods of the resonant converters, however they are more sparse than the frequency modulated ones - another reason I was unsure whether these converters would be applicable.With the forward converter there exists an output filter inductor - which is troublesome in my case, considering the high output voltage, the inductor would be too large, and impractical. I have looked into forward converters, alongside push-pull converters, however literature states that these are used in applications with power levels <500W, rather than >500W where the transformer size becomes a limiting factor. I have looked into interleaving methods for boosting the power capability, but this is typically to share higher levels of input current, which I don't really have - furthermore an additional transformer may be unnecessary.I really value your help with this, if you have any more input it would be greatly appreciated.Thanks!

author: MagicSmoker
author_rank: Super Contributor
timestamp: May 12, 2019, 01:22:48 pm
content: Quote from: state_of_flux on May 11, 2019, 02:43:51 pmAnd the transformer parasitics should concern you the most, because a hard-switched converter with a 10kV secondary has terrible problems with corona discharge damaging insulation and massive phantom losses to stray capacitances due to the high dV/dt. Forming the voltage waveform into a sinusoid and/or lowering the switching frequency will often be necessary.And now that I think about it, the two-switch flyback might be a good choice here. The conventional single-switch flyback tends to be an increasingly bad choice above 200W or so, especially if it has to cover the universal mains input range (e.g. - 85-265VAC), but the two-switch variant doesn't have any problems with leakage which usefully extends its power output much higher, especially for high secondary voltages and/or with a restricted mains input range (or the use of the much hated - and with good reason - switch to select between full wave bridge or voltage doubler mains rectification).Otherwise, you can also consider using a lower voltage secondary and a voltage doubler, quadrupler, Cockcroft-Walton multiplier, etc. One effect of this approach which is usually a disadvantage is that one or more capacitors are inserted in series with the output which limits current, but that might be an advantage here. Also, using fewer turns of thicker wire for the secondary will be easier to wind and can reduce the stray capacitance.Quote from: state_of_flux on May 11, 2019, 02:43:51 pmAye, it might seem like a chicken or the egg problem, but keep in mind that Faraday's equation applies pretty much regardless of topology. I like the version:N = (V*ton) / (dB * Ae)Where N is turns, V is applied primary voltage, ton is the time this voltage is applied in microseconds, dB is the peak flux swing in Teslas and Ae is effective core area in sq. millimeters, because it doesn't require any weird conversions for Gauss or waveform shape, etc. Note that you often end up using the same flux swing regardless of topology once switching frequency exceeds 40kHz or so because core losses dominate (that is, a bridge converter with a bipolar flux swing won't necessarily deliver much more output power than a forward with a unipolar flux swing).Quote from: state_of_flux on May 11, 2019, 02:43:51 pmWell, that's because the transfer function for all resonant converters is non-linear, with the inevitable practical consequence of not being entirely predictable in behavior across all load and input/output conditions. Also, resonant converters tend to be intolerant of either open-circuit loads (series resonant) or short-circuits/overloads (parallel resonant), further restricting their usefulness.Quote from: state_of_flux on May 11, 2019, 02:43:51 pmYes, this is the conventional wisdom that is repeated in every single book I have on SMPS design, yet none of these (highly regarded) authors point out the same problem applies to the transformer secondary. In my practical experience, HV chokes aren't any more troublesome than HV secondaries (why should they be?), but that is not to say that more care isn't required in designing either of them. The real issue is always the voltage between turns, and this is often more difficult to control in a transformer secondary than a choke. That said, if you can relocate the choke so that it handles lower voltage at higher current this is usually preferable.Which reminds me of another topology that would be a good fit - the compound buck current-fed full bridge. This uses a width-modulated buck converter feeding a full bridge run at a fixed 50% duty cycle (actually with a slight overlap of each bridge leg) but without the buck output capacitor; the capacitor after the bridge rectifier on the transformer secondary is reflected back to virtually appear at the junction of the buck and full bridge. This topology is capable of hellacious amounts of power and is an excellent choice for HV outputs, though keep in mind the earlier caveat about hard-switched converters not being the best choice for HV outputs in the first place.Quote from: state_of_flux on May 11, 2019, 02:43:51 pmTransformer size is, indeed, a limiting factor for the flyback, but it is less of an issue for the unipolar forward-based converters unless switching frequency is really low (<25kHz). What really limits power for unipolar topologies are the peak currents in the switches and diodes. For example, the transformer for a 500W half-bridge will be exactly the same size as a 500W full bridge, but the switches in the latter see about half the average current. One might think the full bridge could handle twice the power with the same switches, and it can, but that would require doubling the conductor area of the transformer windings so you end up needing a bigger transformer (in window and/or core area).

author: T3sl4co1l
author_rank: Super Contributor
timestamp: May 12, 2019, 04:43:51 pm
content: Right, at those voltages, you can't meaningfully get a conventional switching supply, you've got to deal with the secondary capacitance. You can run the frequency really low, but that makes the core and capacitors much larger.You canmitigatesecondary capacitance. Multisync CRT monitors did this: the EHT secondary is wound one layer at a time, connected through a diode to the next layer. The wire was extremely fine of course, and still a fairly wide bobbin was necessary (so that the width of a single layer was able to generate several kV, and a huge number of layers and diodes was not necessary). The AC voltage appears across the width of the secondary, the same polarity for all layers -- this nulls out the inter-layer capacitance, allowing sweep rates over 100kHz* (which because of the time required for retrace, means a transformer bandwidth over 1MHz).*Actually, I'm not sure if multisync monitors ever went quite that high. Trinitrons tended to use independent HV supply and deflection. Your average, lower cost (spherical) CRT, up to say 1280x1024 or the like, most likely was combined HV and deflection though.Otherwise, that strongly suggests, either LCC, or a variation on LLC that's set up more like a Tesla coil, i.e., double resonant, but that would be more complicated and harder to control, so I'd suggest not trying that.Fixed frequency: it isn't a deal breaker, but it'll be harder of course. I would suggest, first all, seriously re-examining your constraints, and figuring out if it's really required, or if better filtering for example, will do.For that, I would suggest, inverter (fixed duty and freq), driving the transformer, which has some variable capacitors on the secondary to trim its resonance just below F_sw. A buck converter varies supply voltage to the inverter, thus providing control.The control system is then a normal current-mode controller (buck inductor current), inside an optional voltage control loop (inverter voltage and/or transformer energy -- note the transformer resonance is effectively downshifted and appears as inverter bus voltage, its resonance acting in parallel with the inverter bypass caps), inside a voltage control loop (finally regulating output voltage).Two error amp stages is probably adequate, but beware that it may have to be fairly slow, or need lead-lag components to achieve reasonable phase margin, due to the extra pole that is the transformer resonance. Three stage control may give the benefit of better phase margin and faster overall response (allowing smaller filter caps?).The fixed-frequency inverter should still have some protection features attached, like peak current monitoring, or output current phase (which affects ZVS switching and therefore losses), or even just temperature. Not sure; I'd want to play with it a bit to get a feel for how it might fail, and what signals are appropriate to anticipate that failure.Note that the transformer construction could be adjusted for F_sw, but probably not precisely enough from part to part that you can avoid external tuning. If F_sw can be variable, then tuning won't be required (run the inverter at or near 0 degrees phase shift between output voltage and current), but of course one would then suppose LCC would be acceptable as well.Good luck,Tim

author: state_of_flux
author_rank: Contributor
timestamp: May 13, 2019, 01:25:06 pm
content: MagicSmoker,The input voltage of my application is a DC voltage that varies between 240-300V approximately. I have looked at using the 2 switch variations of the forward and fly-back converters since they allow recycling of leakage energy and clamp the MOSFET stresses to the input voltage - however as you mention yourself the best way to mitigate the issues with HV designs is forming a sinusoid or lowering the switching frequency. The fly-back is still going to present a square waveform to the transformer, and reducing the frequency significantly will increase the transformer/passive component size and weight - disagreeing with my specification of maximum power density.In literature, similar projects to mine do stack multiple secondaries and connect them together in series at the output, which reduces the stray capacitance and allows use of lower voltage rated components - but to what extent can this value be mitigated without affecting circuit performance is the issue.QuoteExactly - I did look into auxiliary circuits to improve the low-load performance of series-resonant circuits, however my converter should operate at full load for a large amount of time, so poorer performance at low-load conditions might not effect overall efficiency too much?QuoteThis sounds like a good possibility. I did look into two-stage conversion topologies with resonant converters at fixed frequency. In this case, could it be possible to modify the full bridge to achieve some soft switching? Possibly the use of snubber circuits across the MOSFET switches?QuoteThis is actually something I hadn't thought about. The half-bridge is definitely not applicable here but I will definitely reconsider other uni-polar topologies if they can reduce transformer size.

author: T3sl4co1l
author_rank: Super Contributor
timestamp: May 13, 2019, 01:36:04 pm
content: Quote from: state_of_flux on May 13, 2019, 01:25:06 pmHmm?They're saying you can double the power with full vs. half-bridge (or vs. push-pull). Obviously the transformer needs to double as well.Unipolar (half wave flyback/forward, and... class E I guess?) have a duty cycle penalty (costs copper) as well as a flux penalty (only half the core's B-H curve is utilized). They're larger for a given frequency.Note that PP has the duty cycle penalty (each half of the primary is used at D = 50%, so sqrt(2)-1 ~= 40% more primary copper is needed), but not the flux penalty. Half bridge has neither, but requires coupling caps or split supplies.Nice thing about LLC is the coupling caps come for free (you need the resonant cap(s) regardless), so half bridge is popular there.Tim

author: state_of_flux
author_rank: Contributor
timestamp: May 13, 2019, 01:41:12 pm
content: Hi T3sl4co1l,QuoteExactly. This is the most important trade-off for me, finding the right switching frequency that reduces the transformer/passive component sizes while ensuring proper operation and utilising the sec. capacitance. The winding configuration of the EHT by Multisync CRT is really interesting - I will look into this. I also looked into the use of planar transformers for high voltage transformer since they allow tighter control of transformer parasitics - but it's unlikely they can be used in voltage outputs at this level without causing significant issues of proximity effect between the layers.QuoteSo you'd suggest LCC is more appropriate than the standard LLC converter? Yes, I'm relatively new to the power electronics field so don't want to make life too difficult for myself - but it's quite a demanding specification.QuoteI actually have recently considered this approach, and it has been used to good effect in similar applications. The pre-regulator might allow me to optimize the resonant solution for a single frequency and I suppose the control might actually end up being less complex than using the single stage resonant converter alone? If you don't mind me asking, what is the benefit of using the variable capacitors to trim to just below resonance rather than just fixing F_sw to be slightly higher in the first place?QuoteThat's interesting - i've never actually heard of transformer external tuning before. I'm unsure what you mean by the last point in regards to the LCC being acceptable - is this because it can provide tuning in some way?

author: state_of_flux
author_rank: Contributor
timestamp: May 13, 2019, 01:45:33 pm
content: QuoteOops, clearly lost in translation there. My bad for misunderstanding.QuoteWith the half bridge converter however, does it not only present half the input voltage to the transformer primary? Meaning an even larger number of turns on the secondary and thus more stray capacitance. As you say I very often see the LLC used in the half-bridge topology in lower power, step-down applications - but rarely see them in step-up, high output voltage applications!

author: T3sl4co1l
author_rank: Super Contributor
timestamp: May 13, 2019, 01:47:16 pm
content: Quote from: state_of_flux on May 13, 2019, 01:41:12 pmYeah, planar transformers have close coupling -- wide, broadside facing traces. No good at HV. Fantastic at low impedances (typically low voltages and high currents, and modest ratios).That's the other thing, high ratios suck, because of trace width and space limits. Besides which, a high ratio winding necessarily has a high impedance, but it can't, because planar.QuoteYeah, because tuning is either setting F_res just so (F_sw stays fixed), or varying F_sw to suit (F_res stays fixed). Or both I suppose, but who would do that.Typical resonant controllers vary output power by controlling F_sw (which, I think you already knew, so this is just going to be a "duh"? It's alright.)Quote from: state_of_flux on May 13, 2019, 01:45:33 pmYa. Half primary turns.Tim

author: state_of_flux
author_rank: Contributor
timestamp: May 13, 2019, 02:01:44 pm
content: QuoteOf course - it's fine, I appreciate your help. It was to my understanding though that if a pre-regulator is used you wouldn't actually need to vary the switching frequency at all - optimizing it for a certain operating point. I'm not sure if you have missed my earlier reply to you? Apologies if you did not.QuotePhew, glad I didn't get this one wrong as well! Otherwise I think even the Beginners forum wouldn't be suitable for me!I'll stick to full bridge then.Thanks Tim.

author: MagicSmoker
author_rank: Super Contributor
timestamp: May 13, 2019, 03:43:48 pm
content: Quote from: state_of_flux on May 13, 2019, 01:25:06 pmThe flyback primary has a square voltage impressed upon it, but the secondary acts as a current source. Remember, the primary and secondary never conduct at the same time in a flyback so the transformer is really a coupled inductor. Consequently, the main design criterion is that Amp*Turns are conserved between primary and secondary. E.g. - if applying 300V to a 10t primary causes current to ramp up from 0A to 10A then upon the primary side switch turning off the secondary diode will attempt to deliver a peak current of 1A (ramping down to 0) for a 100t winding, or 0.1A for 1000t, etc. This is what gives the flyback more flexibility with respect to turns ratio than forward mode converters, though keep in mind that when the switch is on the primary voltage reflects to the secondary according to the turns ratio and the inverse occurs when the diode is on, and this sets the voltages that each device must withstand (plus the input and output voltages, for switch and diode, respectively).Which is sort of a long winded way of saying the flyback - particularly the two-switch variant - is still very much in contention here, especially since it will be far easier (though not easy!) to design compared to any kind of (quasi)resonant topology.Quote from: state_of_flux on May 13, 2019, 01:25:06 pmThis is commonly known as "pi winding" for historical reasons (though I don't have the foggiest clue why) and while it does reduce interwinding capacitance, it tends to greatly increase leakage inductance. Remember the first law of engineering: There Ain't No Such Thing As A Free Lunch!Quote from: state_of_flux on May 13, 2019, 01:25:06 pmIt's not so much a matter of lower efficiency if you open-circuit a series resonant converter or short-circuit a parallel resonant one, it's more a case of instant switch destruction, though it is typically easier to protect the former from abuse than the latter.Quote from: state_of_flux on May 13, 2019, 01:25:06 pmThe bridge switches operate under very benign conditions and it would be best to use IGBTs because they have much lower parasitic capacitances than equivalently powerful MOSFETs and their lazy turn-off is an advantage here as you need overlapping conduction of each bridge leg. It is the buck switch(es) that have to endure exceptionally unfavorable turn-on conditions, and which benefit the most from a lossless snubber (turn-off for the buck switches is relatively benign, however).Quote from: state_of_flux on May 13, 2019, 01:25:06 pmNot the best example I could have used - two bipolar-driven transformer topologies - as the real point I wanted to emphasize is that you can't necessarily double the flux swing in a bipolar topology - and therefore double the power from a given transformer - because iron losses go up exponentially with flux swing, and that giving the forward converter a switch that is just as powerful as two switches in a bridge narrows the gap in maximum power output between them quite a bit.EDIT- clarifications

author: state_of_flux
author_rank: Contributor
timestamp: May 14, 2019, 12:03:01 pm
content: QuoteI suppose then a design trade-off would be selecting the turns ratio as to reduce the parasitic capacitance while also ensuring the voltage blocking capabilities of the MOSFET switches in the primary and the diodes in the secondary don't become unreasonably large? Maybe again the use of voltage multipliers could be adopted? What I like about the fly-back is how suitable it is for multiple outputs and regulation of them - my application can have anywhere from 2+ number of outputs.QuoteBut aren't IGBT's only capable of low switching voltages, say, 30kHz? Would that not mean a significantly larger transformer size, or is there a point in which an increase in frequency no longer has a significant impact on size?The primary design criteria for me is to ensure a small, lightweight solution.So, essentially, incorporating some kind of lossless snubber to the buck-converter for turn-on will allow one to reach a high efficiency solution despite using a hard-switched full-bridge as the main regulator to the HV transformer?QuoteJust so that I understand - what you are saying is that uni-polar topologies can reach similar power levels to bi-polar counterparts just by incorporating a second switch - while reducing the iron losses due to reduced flux swing (since they only occupy one quadrant of the hysteresis loop)?

author: T3sl4co1l
author_rank: Super Contributor
timestamp: May 14, 2019, 12:19:44 pm
content: IGBTs come in different flavors, roughly speaking differentiated by the hFE of the BJT element. Fast ones are capable beyond 200kHz; much of the current flow is through the MOS element, it's more like a MOSFET with a little bit of boost. At turn-off, most of the current drops sharply, then a small tail (a few percent of I_on?) "drools" out as the BJT element continues to turn off (recombination), taking perhaps hundreds of ns to reach full off state (leakage current).Slower ones are limited to lower frequencies, usually the 10s of kHz, even the 1s of kHz for the big modules (100s to k's of amperes, low kVs). These have a smaller fraction due to MOS current flow and more due to BJT amplification. The voltage drop is lower (that's your tradeoff), but the turn-off is not nearly as sharp (maybe it drops sharply by 10-50% at turn-off), and the "drool" is much longer (~µs).Smaller IGBTs always run faster; if you need operation at 100-200kHz say, at high power (industrial application), you may be better off using a stack of TO-247 sized parts in parallel, instead of a module proper. It may be advantageous to operate at lower voltages (600V parts are faster than 1200V+ parts), though that's not always practical for industrial application (where 400/480VAC 3ph input is most convenient).But down at the power levels we're talking here, I don't think it matters. You're mostly paying for the package. What's inside it, doesn't really matter. Might as well go with the MOSFET, and not have to worry about "drool".Tim

author: state_of_flux
author_rank: Contributor
timestamp: May 14, 2019, 01:45:52 pm
content: Thanks Tim.Could you help clarify this point, as I find it an interesting method:QuoteDoes this mean that the primary side would only have a series combination of L and C, then the additional parallel C value to form the LCC resonant tank will be realised in the form of placing variable capacitors in the secondary side between each of the secondary windings. And as I understand it, this is also acting as a way to help tune the transformer, rather than design the transformer according to the switching freq? And tuning its value tunes the transformer by manipulating the resonant frequency of the tank, thus changing the amount of current supplied to the transformer etc? Sorry if I'm sounding a bit dumb, just trying to get my head around it.

author: T3sl4co1l
author_rank: Super Contributor
timestamp: May 14, 2019, 03:44:40 pm
content: The equivalent circuit would be, inverter (constant voltage output), series capacitor, series inductance (including leakage), shunt magnetizing inductance in parallel with secondary self-capacitance.I guess the point of LCC is the magnetizing inductance is large enough to be negligible, and the capacitors and leakage dominate the response. That would suggest an ungapped, high-mu core, which is fine.Primary shunt capacitance wouldn't be needed because the primary will have very little capacitance in comparison to the secondary. From an RF perspective, it's just a magnetic loop to couple energy to the secondary, which is resonant.Also, because secondary capacitance is inevitable, tuning the primary would probably be folly -- either you have so much leakage that you end up with a double-tuned network anyway, or the leakage must be extremely small so that Cpri acts in parallel with Csec. But that's going to be impossible with the number of turns and thickness of insulation required on the secondary.So, the capacitor just goes on the secondary instead, which is fine, and it can be tuned there similarly.Uh, the one catch, of course... finding a 20kV trimmer.Probably better to do it with "gimmicks", e.g., pours on the PCB that can be cut apart or jumpered together. (FR-4 is a terrible capacitor, not so much because of losses at this frequency, but because of tempco -- it might be good enough, or it might exacerbate whatever tempco the transformer has. Or maybe it cancels out and is actually best. No idea, you'd have to test.)Also in your favor, the Q factor is supposed to be quite low for resonant supplies, so tuning shouldn't have to bethattight. Probably tempco doesn't matter, and swapping out a few leaded ceramic caps will do.Incidentally, if stacked secondaries are used, you only need to tune one or a few of them -- again, they act in parallel at AC, in series at DC. As long as the leakage between sections isn't huge (again, same argument as leakage to the primary), they'll behave. And that means you only need, say, a 5kV cap, instead of the full 20kV or whatever.Tim

author: state_of_flux
author_rank: Contributor
timestamp: May 14, 2019, 04:12:38 pm
content: Seems quite complicated.I'm quite a newbie so honestly its quite overwhelming but I appreciate your help regardless. I suppose if one was to use voltage multipliers at the output also, this would further reduce the value of variable capacitance voltage that would be needed to be placed on the secondary, since for example voltage doublers share the output voltage value across them both (Vo/2)? If it did turn out only the first secondary winding needed to be tuned, would it then be that the other secondaries would just have standard HV capacitors and wouldn't need to be trimmers?So given what we have discussed in this thread, do you suppose the buck converter fed LCC is the best way forward topologically for the design I have outlined? You've definitely answered my question in regards to whether LCC or LLC is better for high output voltage, but do any other possible approaches spring to mind at all?Cheers

author: MagicSmoker
author_rank: Super Contributor
timestamp: May 15, 2019, 02:33:42 pm
content: Quote from: state_of_flux on May 14, 2019, 12:03:01 pmYes, though it is a seesaw type of tradeoff: a higher turns ratio (from primary to secondary; ie - step-up) reduces the blocking voltage requirement of the switch (though see caveat below on constraints with the two-switch variant) but increases it for the diode, and vice versa.With the two-switch variant, however, the switches never see a voltage higher than the incoming supply (plus two diode drops) and leakage spikes are non-existent (assuming the clamp diodes are sufficiently fast). This sounds great - and it is - but that means the turns ratio must be set such that the clamp diodes do not conduct during the period when the output diodes are conducting, only at the very beginning of switch turn-off, to return leakage energy back to the supply. In other words, the voltage reflected back to the primary when the secondary is conducting must be less than supply voltage, which means a higher minimum turns ratio. For example, with a 335V nominal supply and a 10kV maximum output the turns ratio must be at least 30. That sets the minimum turns ratio, but what sets the maximum? Well, the higher the turns ratio the higher the voltage the output diodes have to withstand during the switch on time - that is a clear disadvantage of this configuration with HV outputs as now the output diodes have to withstand at least double the output voltage. Also, a higher turns ratio results in a shorter flyback period, which means a higher peak current in the secondary diodes, so just make sure this peak current is acceptable. Again, this is something which is unlikely to be a deal-breaker with a low voltage output, but high speed diodes rated for >10kV and more than a few hundred mA are few and far between. In fact, the only one I can think of off the top of my head is the somewhat antiquated (likely obsolete) 2CL2FM which (IIRC) is rated for 100mA to 200mA average and 20kV reverse voltage with a 100ns reverse recovery time. Finally, the higher the high turns ratio (in either direction) the higher the leakage inductance, usually, (and the higher the losses from proximity effect, which, fortunately, does not apply to the flyback). While the clamp diodes in the two-switch variant do return all the energy stored in the leakage inductance back to the supply, while they are doing so the output diodes can't deliver energy to the load... So, a lower turns ratio secondary feeding a half-wave doubler/tripler/quadrupler could be highly advantageous here for that reason, as well.And if you put all of these conflicting issues together you start to see why I keep recommending a flyback with a voltage multiplier, despite it not appearing to be the best choice at first glance. Yes, a properly designed (and protected) LCC or other resonant converter would achieve a higher efficiency and likely take up less space, but forget this being the deep end of the pool; at that point you are swimming with the sharks out in open blue water. If you have sufficient dedication and budget (in both time and money) to dick around with this, then by all means attempt a resonant converter design - it will be better for the job - but if the real world has placed practical restrictions on you then, no, none of the exotic topologies are really appropriate. Note that I am not saying you can't or won't be able to get a more exotic topology working - who am I to judge what you are capable of? - I'm just saying that it will be a much more difficult task, relatively speaking.Quote from: state_of_flux on May 14, 2019, 12:03:01 pmHV and "small, lightweight" don't really go together. Minimizing the size requires vacuum impregnation and potting to prevent failure from corona or outright arcing, while minimizing the weight requires avoiding potting and relying on generous spacing between components to achieve the withstand voltage.And you won't be running nearly as high as a switching frequency as you appear to think is possible - after all, a mere 10pF of stray capacitance has an impedance of 160k at 100kHz, so a 10kV square wave applied across this will result in 63mA of current. Whether that current is flowing across the junction of a diode that is supposed to be off, or across a transformer secondary, it amounts to a pure loss...Quote from: state_of_flux on May 14, 2019, 12:03:01 pmThe bridge in a buck current-fed converter doesn't really operate under hard-switched conditions - the overlapping on time ensures that turn-on is at 0V, and the buck inductor limits current rise at that time, too. The only real downside is that the last transistor to turn off in each leg experiences an overvoltage spike, but that can be tamed with a snubber or a clamp connected across the buck switch(es) and inductor.Quote from: state_of_flux on May 14, 2019, 12:03:01 pmSorta not really. I was mainly saying that there is not such a huge advantage in power output between a bridge and a forward converter once the switching frequency is high enough and the forward has a switch with twice the average current rating as any single switch in a bridge. What is "high enough" for the switching frequency? Well, that very much depends on a number of factors, obviously, but in my experience, starting around 50kHz and 400W of power the total flux swing ends up being limited to the same value regardless of whether operation is confined to one quadrant of the B-H curve or two. A bridge converter will still deliver more power from a given core than a forward (and much more than a flyback), it's just that the real advantage of the bridge is splitting the power up among more devices (and being able to use full wave rectification on the secondary). Consequently, I would be more inclined to go with a two-switch forward even as high as 600W, but as mentioned earlier, this applies mainly for low output voltages, as for HV outputs the flyback starts to look better, despite the higher peak currents, owing to more flexibility in transformer turns ratio and no need to worry about proximity losses (which to minimize requires interleaving the primary and secondary windings).

author: T3sl4co1l
author_rank: Super Contributor
timestamp: May 15, 2019, 03:12:07 pm
content: Quote from: MagicSmoker on May 15, 2019, 02:33:42 pmHm? Explain?Proximity effect occurs in any multilayer section. It's simply the image current of one layer induced upon the adjacent layer, that happens to already be carrying the same current in the same direction.It probably isn't important at these currents and frequencies -- the wire used will be quite small.It's like how the smallest wire I have in stock is 37 AWG, so when I want to wind something for high voltage, well, it's automatically capable of ~30mA continuously, even if I only needed a few mA...So, which, actually... thanks, I never thought about this before but it's quite true:In any flyback transformer, with multiple layers of windings,regardless of interleave,proximity effect occurs.This is simply because the primary and secondary currents are not coincident, so they can't cancel out their proximity. (Whereas you can do this in a forward converter with interleaved single layer sections.)So flybacks with many layers are somewhat prohibitive, and one or few layers are preferred, which means wider bobbins should also be preferred (i.e., to maximize turns/layer). Cool!(Interleave is still critical for leakage, however!)QuoteIn any case, the 2-switch discussion is spot on. Also, the secondary peak voltage rating is the same (total) whether the secondary is whole, or broken into sections with a diode each.Incidentally, HV diodes are all just stacks of lower voltage diodes -- I don't think they can make high speed (<= 250ns or so?) diodes over 1.5kV or so. (In fact, an industrial "hockey puck" rated for 6kV is "high speed" if it has a recovery of 2us!) So you see V_F climb -- multiple junctions in series.The trick is, they're matched diodes, and packaged together, so recovery and capacitance are equal. This is a lot harder to pull off with discrete diodes*, so there is value in the high voltage types.*What happens is, one recovers first, and gets all the voltage drop and immediately goes into avalanche, burning a huge amount of power. Then the next one recovers, the total voltage drop rises and not as much power is burned, and so on until the total avalanche rating exceeds the applied (reverse) voltage, and conduction stops. And it's a runaway condition, because recovery gets slower at higher temperature.At least it's not as brutal as with a forward converter, where the reverse current is switched in -- a flyback in DCM has soft recovery, with dI/dt set by the transformer inductance.Actually, an even more subtle point is that recovery itself is a sort of dynamic avalanche condition: effectively, the off-state (blocking) voltage of the diode increases over time, eventually reaching its nominal rating. This probably happens simultaneously for all diodes in the stack, but the result is still that one diode dissipates much more recovery loss than all the others.But anyway -- if 20kV diodes are unobtanium, at least the secondary can be split into sections, and as many 3 to 10kV diodes can be used, which are hopefully more obtanium!QuoteAlso -- note that a multiplier needs a different control circuit than a bog-standard (current mode) flyback. The multiplier draws current during switch-on, so either a current-limiting switch is needed, or something that acts in the same way. It may be adequate, for example, to use a normal peak-current-mode controller, but increase the transformer's leakage, so that the switch turns on for a moment, charges the multiplier through the leakage inductance, then turns off. (The increased leakage would suck for a one-switch converter, but would be very much in favor of the two-switch or full-bridge version.) The operational result will be, lower maximum power output at low output voltages, because the switch turns off much sooner than it would otherwise. If you don't need a wide output voltage range, and don't need a lot of startup current, that should be quite feasible.A current limiting switch (i.e., a linear device) is probably completely out because of heat and size.Proof that it can be done, though:https://www.seventransistorlabs.com/Images/Deadbug_Sch.pngThis is hardly a watt, and the switching transistor can get quite toasty under some conditions, so it's hard to recommend at any kind of scale!QuoteAnd speaking of budget -- do consider hiring a consultant to help guide you. Or, heck, just contract out the whole thing, if it just needs to be done -- give or take how many lessons you want to learn from the project as well.Which, on that note, I'm pretty busy right now actually, but if your timeline extends to late summer I'd be glad to help. Otherwise, there may be high voltage / switching professionals here who are available; doesn't hurt to ask.QuoteWell..... that's the whole point of the resonant converter, right?It does tend to be loss, in a traditional flyback converter that needs wide bandwidth and can only discard energy that gets trapped in reactances.It does seem unlikely to construct a transformer with bandwidth of multiple MHz (that would make a 20-100kHz flyback feasible), so that is a point strongly in favor of resonant.The tradeoff is then, a flyback that's easy to design but has terrible efficiency, or a resonant that's hard to design but easily gets good efficiency.Tim

author: MagicSmoker
author_rank: Super Contributor
timestamp: May 15, 2019, 04:49:48 pm
content: Quote from: T3sl4co1l on May 15, 2019, 03:12:07 pmYes, an important point I failed to make - that proximity effect applies whenever a winding has more than one layer because the fluxes from each layer add together, so it was misleading of me to say that proximity losses aren't a problem at all in the flyback, it's just that you can't do anything about them except minimize the layer count and that is true of any magnetic component that experiences a significant flux swing. The one thing the flyback has going for it compared to forward mode converters is that the current waveform is a ramp and proximity losses are also proportional to the rate of change in the current (ie - flux).Quote from: T3sl4co1l on May 15, 2019, 03:12:07 pmYep - the Vf spec for the 2CL2FM I mentioned earlier is something like 15-18V, but the junction capacitance is probably so low it is unmeasurable.Quote from: T3sl4co1l on May 15, 2019, 03:12:07 pmIt's not as much of a problem as you might think AS LONG AS you always use a static balancing resistor and dynamic balancing RC network across each diode in series. Incidentally, the design of the RC network for a series string of diodes is much the same as for a dV/dt snubber for a thyristor.Quote from: T3sl4co1l on May 15, 2019, 03:12:07 pmYes, another excellent point - I know this trick has been done when you need to supply precisely stepped voltages to the anodes (dynodes) of a photomultiplier tube, but that hardly qualifies as a load; probably not nearly as practical if you need to draw more than a few tens of mA. Standard peak current mode controllers handle it just fine, btw (again, for relatively low output powers...)Quote from: T3sl4co1l on May 15, 2019, 03:12:07 pmThat's the rub - the problems with getting a resonant converter to work may very well be so difficult for the first time designer that not much might be learned in the process. As some old wag once quipped - multiplication doesn't make any sense until you learn addition.Quote from: T3sl4co1l on May 15, 2019, 03:12:07 pmBingo. A flyback run at as low a switching frequency as you can get away space/volume permitting, will be much, much easier to get working.

author: MagicSmoker
author_rank: Super Contributor
timestamp: May 16, 2019, 12:14:52 pm
content: Quote from: MagicSmoker on May 15, 2019, 04:49:48 pmThe above comment got me to thinking so I pulled out an LTSpice circuit I drew up awhile ago for a two-switch flyback and modified the transformer parameters so it would produce 5kV/0.1A which could be fed to a doubler (Villard and C-W were tried). I didn't bother finding a proper HV diode model so I just used the default diode (infinite blocking voltage! / infinitely fast!) with a few puff of simulated junction capacitance.* Unsurprisingly, the leakage inductance parameter needed to be set to something realistically high** for this to even work, and trying to pull more than a few mA caused the voltage to collapse to the peak output of the transformer. And note that this is without any parallel capacitance across the transformer secondary, which would only make things worse.So, I rescind my recommendation to use a flyback with a lower voltage secondary feeding a voltage multiplier - it might be fine for biasing the anode(s) on a CRT or photomultiplier, but not if any kind of real current must be drawn.I'll attach the .asc file for the flyback with a conventional rectifier output if anyone wants to play with it. The transformer core is a Magnetics, Inc. EC 70 shape in P material with a 0.5mm gap and a 1:30 turns ratio and if not optimal for the application, is reasonably close. Other caveats: I did not optimize the feedback compensation; the transformer secondary still doesn't have a parallel capacitance specified; the primary side RC damper to suppress ringing is not accurately specified since the secondary diodes and transformer parallel capacitance aren't specified.* - LTSpice seems to have a hard time if ideal diodes are used in power circuits; putting a small capacitance across them is the trick that makes it happy.** - e.g., 0.965 gives a leakage of about 7%, which is very good, but not impossible to achieve, for a HV transformer with a 30:1 turns ratio.

author: T3sl4co1l
author_rank: Super Contributor
timestamp: May 16, 2019, 12:49:46 pm
content: Oh neat, haven't seen a GDT drive like that before. Makes sense. Think I'd put a diode across Q4 too, but it's not required.Tim

author: state_of_flux
author_rank: Contributor
timestamp: May 16, 2019, 03:43:14 pm
content: Thanks guys.QuoteSo, from re-reading this thread, that only really leaves me with the buck current-fed full bridge, or the LCC running at fixed frequency, fixed duty, with a buck pre-regulator to achieve control? This is a project that I plan on running into next year so I do have time to attempt a more complex solution if that is the case and it doesn't seem as though I have many more options available.MagicSmoker, I have one point in regards to a previous point you made:QuoteIsn't it the case though, with the LCC converter, that it is naturally protected in overloading and short-circuit conditions due to the parallel capacitor? In my application, short-circuits can and do occur - so again this could point towards the use of the LCC.One issue I do have with the LCC/LLC resonant converters is their controllability - it is to my understanding that both these multi-resonant converters have limited usefulness due gain variations and chaotically moving poles and zeroes in their dynamic power transfer function. I've seen Texas Instruments use what's called Hybrid Hysteretic Control to overcome this,https://www.mouser.co.uk/new/Texas-Instruments/ti-ucc256303-controller/.It says that the system is always stable with proper frequency compensation, but again in my case where this is most likely not okay - would it make these devices inapplicable and thus meaning a terrible control characteristic regardless of choosing LLC and LCC? Or could this be alleviated through the use of a pre-regulating buck converter again?The load of my converter is typically pulsed on/off with a pre-determined duty cycle, and the regulation is quite stringent - so I believe this to be an important point I neglected to mention.Also, T3sl4co1l, I have dropped you a quick message in regards to consulting, in case I am still banging my head against the wall in late Summer.

author: MagicSmoker
author_rank: Super Contributor
timestamp: May 16, 2019, 06:17:37 pm
content: Quote from: T3sl4co1l on May 16, 2019, 12:49:46 pmFeel free to steal it (I suspect its been reinvented plenty of times, anyway)! It's what I consider a "cheap and cheerful" approach but which does a good job of clamping the MOSFET gates in the face of huge dV/dt when they are supposed to be off without requiring the usual mess of parts on the GDT secondaries. It's pulse width fidelity is a bit sloppy, however, so not recommended for bridges and the like.EDIT - I forgot the best part about it: you can use a more common 1:1:1 GDT, rather than a 1:2:2

author: MagicSmoker
author_rank: Super Contributor
timestamp: May 16, 2019, 08:17:04 pm
content: Quote from: state_of_flux on May 16, 2019, 03:43:14 pmThe LLC isn't very accommodating of a wide range of input or output voltage, so it's really off the table completely. In fact, the only times I've ever seen it used commercially is when there is a PFC stage in front of it. Also, it's running neck-and-neck with the PS-FB as the most difficult converter to design. The buck current-fed full bridge is a good choice mainly because: a) the bridge switches have a relatively easy life; b) you can control the output damn near all the way down to 0V; c) it is highly tolerant of short circuits and, of course, doesn't care about open loads; d) the buck converter is the easiest of them all to stabilize. It's main downside - and it is a doozy - is that it's still a hard-switched converter so stray capacitances, in particular, are going to be an immense source of losses. Also keep in mind that stray capacitance on the secondary of a step up transformer will be multiplied by the square of the step-up ratio. E.g., a 30x step-up transformer will reflect the capacitance at its secondary back to primary as 900x higher. Conversely, stray inductance on the secondary will be divided by the square of the step-up ratio, hence it is less of a concern with step up transformers (at least from the perspective of the primary).Consequently, the LCC would be the much better choice between LLC and LCC if for no other reason than it automatically incorporates the stray capacitances into the shunt capacitor that goes across the transformer primary, but it will still be very difficult to get working, especially since the LCC converter has "voltage gain" and it will be destroyed if the switching frequency drops down too close to the resonance point of the series LC network (the shunt capacitance across the transformer is typically smaller than the series capacitance, though see above in which the reflected capacitance might very well be much bigger than expected). Also, the switching frequency of the LCC spans a wider range, typically, than the LLC (especially if you also need to vary the output voltage over a wide range).Quote from: state_of_flux on May 16, 2019, 03:43:14 pmAh, well, you have plenty of time to blow things up repeatedly; so long as you have the budget!Quote from: state_of_flux on May 16, 2019, 03:43:14 pmNo, the shunt capacitor actually makes the LCC more prone to failure from a shorted load, though as mentioned above, this capacitor is typically smaller than the series capacitor and so doesn't dominate the converter's behavior. In fact, a properly designed LCC can tolerate a fairly wide load range (I hesitate to say from open to short, but on paper it is capable of such). Look up a paper from TI titled slup376 for a better explanation and comparison of the LCC and LLC.Quote from: state_of_flux on May 16, 2019, 03:43:14 pmSorta - it is probably more correct to say that the transfer function of resonant and multi-resonant converters is non-linear and often has no formal solution at certain combinations of line, load, transformer ratio, etc. And by that I mean the converter will fail to regulate correctly, latched itself at zero output, or blow itself up.Quote from: state_of_flux on May 16, 2019, 03:43:14 pmI didn't read the link, but understand that the usual definition of hysteretic (or the more colorful, "bang-bang") control is to turn the switch on when the output drops a certain amount below the setpoint and turn the switch off when it rises a certain amount above it; the width of this error band being the hysteresis. Hysteretic control is woefully underappreciated, in my opinion, because it needs no frequency compensation and has the fastest transient response possible. It's downsides are variable frequency operation and a need to balance the upper limit of such against the tightness of the setpoint regulation (whether the setpoint is for output voltage or current or power).Quote from: state_of_flux on May 16, 2019, 03:43:14 pm*groan* So you need 0-10kV output and up to 50mA, tight voltage regulation, tolerance of shorted and open loads, fixed frequency, compact size, high efficiency... anything else I forgot?

author: T3sl4co1l
author_rank: Super Contributor
timestamp: May 17, 2019, 01:43:18 am
content: Quote from: state_of_flux on May 16, 2019, 03:43:14 pmGot it, will check back in laterTim