Cleaning up a very noisy flyback DC converter.  - Page 1
Forum section: electronics / projects

author: Fflint
author_rank: Frequent Contributor
timestamp: February 25, 2022, 01:44:21 pm
content: I have a cheap dc-dc flyback high voltage converter I would like to use in a project that requires as low EMI as possible. I'm looking for advice and any written material that will help me modify it to accomplish that.I also built an enclosure shielded with copper, but removing the noise in the source is much better than trying to shield it.It appears the noise is switching ringing that happens on the primary side. I think so because the primary side voltages are somewhat similar to the noise waveform and the noise is much stronger when there is a piece of wire attached to the transformer primary.So I probably need an RC snubber. If anyone could suggest any resources that describe how to choose the component values to limit EMI (rather than just protect the switch from overvoltage) that would be great.I show the noise waveform below (cyan). Also there is the transformer primary (purple) and secondary(yellow).I'm mostly interested in cleaning up the noise in the HF frequencies band (3-30Mhz). This is a spectrogram of the noise (5mhz per division).

author: thm_w
author_rank: Super Contributor
timestamp: February 25, 2022, 11:18:05 pm
content: Snubber, gate resistance, transformer shielding, common mode LC filter on input.Is there no way you can switch to another supply?https://www.eevblog.com/forum/projects/rc-snubber-network-design/https://www.cde.com/resources/technical-papers/design.pdf

author: T3sl4co1l
author_rank: Super Contributor
timestamp: February 26, 2022, 01:44:17 am
content: Have you traced the circuit yet?The risetime looksponderouslyslow. Is this also at light load, or..?If the slow risetime is due to snubber capacitance, it appears there is considerable ringing associated with that loop. It will have to be moved closer to the switch, or the layout scrapped and redone entirely.The ringing during switch-on is at the secondary, and can be damped with an R+C across the diode. The ringing during switch-off is at the primary, and likewise, or sometimes an RCD snubber (of the dV/dt or peak-clamping type), and can include a second R+C to dampen the free ringdown period (low frequency ringing).The transformer can also be wound to improve shielding. It is not just a ball of wire, but an entire electromagnetic component all itself -- perhaps obvious on reflection, but the point is it's a far richer object than a mere ratio.What level of noise (amplitude vs. frequency) are you looking for?Another related thread:https://www.eevblog.com/forum/projects/snubber-3-different-230v-ac-motors-in-a-record-cleaning-machine/specifically about mains snubbers, but it's still just an RLC circuit. For SMPS, you will have a stronger concern about power dissipation, i.e., use the smallest C while still giving acceptable dampening.Tim

author: jonpaul
author_rank: Super Contributor
timestamp: February 26, 2022, 10:13:16 am
content: Cleanup of cheap poorly designed junk (Chinese) is a waste of time.These are just bandaids on a cancer:Suggest to throw it away and Buy or build a decent converter,Jon

author: Fflint
author_rank: Frequent Contributor
timestamp: February 26, 2022, 02:45:45 pm
content: Thank you for the advice and links sent so far.I haven't traced the circuit, but it appears a very simple one. It uses a single low voltage mosfet as a switch and a transformer that looks like it came from a computer power supply (center tapped on secondary).There is no snubber on primary, the positive supply simply goes through the transformer primary, then through the mosfet and a current measuring shunt to ground.There is also no snubbing of any kind on the secondary. Just fast diodes for rectification and 10uF caps to filter output ripple.The circuit uses a separate comparator Ic to set voltage and as overcurrent protection. The main control chip is UC3843A.As my target noise figure, I haven't got a numeric answer ready. There is a receive RF coax (rg58) on the other side of a 3mm steel sheet plate. Currently the noise is so bad it is practically unusable. Based on the S meter reading I would like it to be at least 42dB less, ideally 54dB less in any given 2.6kHz passband within HF(1-30mhz). Perhaps the SA screen gives you an idea how noisy it is now.I have a bunch of them and if fixing it means rewinding the transformers to add shielding between primary and secondary, adding capacitors between the primary winding ground and secondary center tap, and adding two sets of snubbers that doesn't seem like that much work. If that fixes it I'll be happy, but the more I read on the subject the less I believe it is an achievable goal. I didn't realise denoising flyback converters is so difficult.An alternative is to build a linear power supply. I'm pursuing both options at the moment.

author: NiHaoMike
author_rank: Super Contributor
timestamp: February 26, 2022, 03:40:18 pm
content: How much voltage is "high voltage"? And what is it used for and how much current? Might it be possible to only run the converter when good noise performance is not important and use a capacitor to hold up the voltage when low noise is important?

author: Fflint
author_rank: Frequent Contributor
timestamp: February 27, 2022, 12:50:23 am
content: Quote from: NiHaoMike on February 26, 2022, 03:40:18 pmHigh voltage is by definition over 42V :-)Unfortunately it isn't possible to do what you're suggesting. The converter outputs max 200mA and I need 30mA on average.Regarding voltage and application what difference does it make? Regardless of voltage, current and application I suppose de-noising techniques for a flyback converter are similar. If there are any publications on the subject please post links, book titles etc.I certainly don't want this thread to devolve into discussion of alternative power supplies to use in my application. That's why I didn't mention it. Also I have additional uses in mind for those boards if I can make them RF quiet.If you are curious my current use is 380V exactly, 30mA on average, spiking to up to 300mA at times (taken care of by a capacitor bank). The PS is used as a screen supply in a HF tetrode power amplifier(with additional negative current sinking and overcurrent protection). I know all alternative ways people use to achieve the same goal (linear supplies from another voltage source, huge power resistor and string of zeners from b+ etc), but I would like to use a specific SMPS board I have lots of if possible.

author: T3sl4co1l
author_rank: Super Contributor
timestamp: February 27, 2022, 02:34:46 am
content: Well, nothing inherently bad about the UC3843, but will have to see the rest of the components and layout. Sounds like they probably don't have any placement options for snubbers so you'll have to tack that all on yourself; rewinding the transformer is a PITA, and, so I guess this is an isolated supply as well -- you'll need line filters for that. Which actually, can be input or output, and I suppose with a high voltage output you might be better off with the latter (still needs differential mode filtering at the input!). But it's a small difference, basically saving on wire size, you put the CMC on the lower current/higher voltage side.Or, sometimes you need CMCs on both, because there's too much noise current into ground (usually heatsinks and shields).The biggest difficulty in the transformer design is, consider the capacitance between layers in the winding. It's a capacity of, whatever the overlapping area is, the tape thickness, and dielectric constant. It's not perfectly that capacitance from terminal to terminal, it's distributed, so it averages out over the voltages present on that layer -- each turn sees a little more or a little less voltage than the last, and so on, right? -- so we can draw up an equivalent that's somewhat less than that, in that case.The best paradigm for transformer design, is transmission line techniques. For example, if you wanted a 1:1 flyback, you could use a single layer each, of, however many turns are needed to handle the voltage on the given core (which hopefully all fits in just the one layer--), and just wind primary then secondary like that. Then the start of the primary we can set to DC_P (which is to say, RF GND), the finish to the switching transistor, and likewise the start of the secondary to output GND and the finish to the rectifier. Consider the pairwise path the wires take around the core: both start ends come into the bobbin, and go around in parallel, as if twin lead cable, wound edgewise. Thus we have a consistent transmission line and impedance and all of that, probably around 100 ohms impedance, and thus LL and Cp corresponding to Zo and c_0 and the wire length of the winding. No need to measure or guess, the calculation is quite close to actual here (say 20%?).And since the hot end and cold end of both windings are in the same places, common mode is very small, and also there's very little effect on the switching waveform -- the interwinding capacitance isn't involved so it can run quite sharp / fast as well. Imagine tying the cold ends together: the voltage difference on the hot end would be nearly identical, save for leakage inductance, and so only that current (the switch from primary to secondary current flow) returns through that cold-end tie. Which is to say, the common mode noise source here.For 380V output, you probably need a 1:2 ratio. This can be done by winding four strands together, connecting two in series for the secondary, and two in parallel for the primary. Which is fine to do directly with TIW, or needs to be done bifilar (two layers, tape insulation between), or maybe with four layers interleaved and then wired in this way. Now you've got the transmission line impedance of the primary start half over the secondary (the cold end, so, 0-0.5 * switch node voltage is imposed on the interwinding capacitance), and the finish half (the hot end, 0.5-1 * switch node voltage). For an average of half. Which isn't great.You can also structure it as a 1:1 isolation, paired with a 1:2 stepup, on a shared core. The 1:1 section is just overlapping primary and secondary windings, as with the earlier case. Then the 1:2 is done by winding a twisted pair (transmission line!) on top of the secondary and connecting it in series with the base winding. That is, the start end of the pair is wired to the terminals of the secondary winding; the finish end of the pair is "stacked on top" of the secondary winding and that's your total output. Which seems wasteful, because, the start end of the winding proper, is identical to the start end of the twisted pair, right? Indeed it is, but minus one catch: the flat winding acts in part as a shield to the primary. That said, if we leave it in place, then wind the extension on top of the secondary (so the sec 1st layer goes e.g. left to right, then we jog the wire back to the left and continue winding), that's not very different, and doesn't require any faffing with twisted pair.Notice the above is different from the usual way two layers would be wound: e.g. 1st layer left to right, 2nd layer right to left. In that case, the secondary looks like a transmission line that is short-circuited at the right end (in the middle of the actual winding), then wound to the left. In a somewhat poor winding, because the helices they trace out are oppositely handed: they cross over each other, making a weird path rather than a smooth parallel-wire transmission line. The weirdness averages out for frequencies much less than a turn length though, just giving, I think, an overall somewhat higher impedance. Anyway, the base half of that winding is coupled to the primary, and the top half isn't. So you get some weird transmission line stub modes in this way, due to that foldback path. You still get an effect of doubled-up TLs in the left-right, left-right windup, you can't avoid the extra delays (unless you're willing to match the delays, with a delay line of the same winding length but making no loops around the core -- but that's not gonna be worth the trouble), but it's going to be better than the foldback path I think.You can also just do dual secondaries and stack them. This has the best value in terms of EMI, when it's a viable option. Consider a three layer 1:1:1 stack, primary in the middle, each section a single full-width layer. Put a diode on each secondary half, and connect them in series. You can have the midpoint of the secondaries bypassed as RF GND. With the diodes on the same ("hot") ends, you get the same situation as earlier, with only the current flowing to leakage inductance as common mode, no switching node coupled between layers.This is a version of what CRT flyback transformers did, back in the day: to support fast scan rates (from basic 31kHz VGA, to over 100kHz at high resolutions), the transformer had to have extremely low capacitance, while still offering a huge stepup ratio to derive the high voltage (20-30kV). They used a relatively wide winding, so that each secondary could be a single layer, and put a diode at the end of each layer. Then just use enough stages to get the required voltage, and that's your high voltage supply. The interwinding capacitance sees no AC voltage drop, because the secondaries are all doing the same thing, and instead serves as bypass between stages. Also the diodes only have to be rated maybe 10kV each, not the full 30+.There's also the consideration for just shoving in a shield. Put down a layer of foil, and leave it insulated so it's a slitted cylindrical shell, not a shorted turn. Connect to the low end (GND or whatever) of the respective side being shielded, and there you go. The downsides are an increase in winding capacitance and leakage inductance: effectively the shield acts as another conductor in a now-wider transmission line, so the TL impedance of primary over shield for example, adds to secondary over shield. So maybe it's double, like 200 ohms characteristic impedance.And as for impedance -- at your modest currents, these are fine. You generally want Zo ~ Vsw/Isw, that is, the peak voltage and current seen at the switch. In practice it doesn't matter much, you just don't want to be so far out (>10x?) that parasitics compromise the thing. It's compounded by winding length, so you generally want to keep that short as well. Which can encourage a fatter/thicker core like RM or P, over a thinner core like EE or ETD. Conversely, a thinner, taller core (wider winding area) may be beneficial for getting single layer windings.There's also compensation windings (winding some turns in reverse, or "overwind", and not connecting to it in circuit, just using it for the capacitance to cancel out with the rest of the winding), etc.Or resonant supplies, which get quite attractive for situations where winding or device capacitances are relatively high (so, particularly for high voltage windings, or high operating frequency at high efficiency). Which can range from quasi-resonant (basically flyback, but wait until the flyback pulse swings back down near zero to turn on again -- less voltage across the switch at turn-on), to full resonant (output responds to change in input frequency/duty cycle perhaps over several cycles) which is obviously a bit more work to design. Both of these are probably out of scope, of course. (Might be a way to hack QR for a '3483, but other than that..)Tim

author: Fflint
author_rank: Frequent Contributor
timestamp: March 03, 2022, 12:48:31 pm
content: Thank you for this it is very useful. I'm currently working on a linear supply for the use I had in mind for that DC converter, but I have few of them so I'll come back to trying to clean them up sooner or later.