FFT Spectrum Analysis Reviewed - Page 1
Forum section: electronics / testgear

author: Performa01
author_rank: Super Contributor
timestamp: April 17, 2016, 02:32:26 pm
content: FFT appears to be a strongly demanded feature in DSOs and there is an ongoing discussion how it compares to a ‘real’ spectrum analyzer and particularly whether it can be useful with just 8 bits resolution in a standard DSO.Traditionally, there are two camps of totally opposite opinions:8 bits might be low resolution and have a poor dynamic range in itself, but this can be vastly improved by certain resolution enhancement techniques. In particular, the FFT in itself acts as multitude of band pass filters (number of ‘bins’) which leads to a dramatic noise reduction for each individual bin. In short, we just need a high number of FFT sample points in order to achieve a decent dynamic range. The catchword here is ‘process gain’.8 bits are low resolution and the dynamic range of even an ideal 8-bit A/D converter is only 48dB. This is just the laws of physics and we cannot get anywhere, no matter what mathematical tricks are used for signal processing. In short: crap in -> crap out.Even though I’ve always held the view that the latter position would be the much safer bet, it is still obvious that the truth would be somewhere in between these two extreme positions.People advocating the ‘process gain’ as the solution for the limitations of an 8-bit system sometimes even pull out some textbook article to back their claims. The catch is – as always – that it is just theory. As we all know, in theory there is no difference between theory and practice, but in practice there is…Of course, the theory is right if we have an ideal system, which cannot exist in real world, plus there’s even more requirements…If we actually had an ideal 8-bit ADC, it should be obvious that a signal with a peak-peak amplitude of less than 1 LSB just cannot produce any change in the output code of the ADC. Consequently, any signal lower than some 48dBfs (decibel below full scale) would just be suppressed. That means, we can well have a very high SNR (signal to noise ratio) due to ‘process gain’ but still the usable dynamic range is limited to 48dB.To overcome this, we need to add some dither – usually white noise – and we will inevitably have to use averaging to hopefully get some meaningful readings for the spectral components below -48dBfs whose readings are now very unstable since they are riding on top of the dither.And here the troubles start already. Who actually adds dither? It is not so easy either. For a 50 ohms input, we can just use an off the shelf resistive power combiner and a suitable wideband noise source, but what if we want to preserve one of the very few advantages that the general purpose DSO can have over a genuine SA, namely the high impedance input? We’d need to construct a power combiner with two 1Mohms ports for the scope and the probe and one 50 ohms port for the noise source. In any case this means 6dB attenuation, thus increasing the noise figure and lowering the sensitivity by this same figure.On the topic of noise figure, the amplitude of the dither should be reasonably accurate. If it is too low, the dynamic range will not be extended by much – if at all. If it is too high, it will decrease the SNR, but then again, this is something where a narrow ‘bin bandwidth’, i.e. a high number of FFT points actually helps.If the DSO front end and/or the signal source are very noisy, we might get the dither for free. But this is rather ill-defined. The scope frontend might contribute enough noise at the high vertical sensitivity settings such as 5mV/div or lower, but is not guaranteed to do so at gain settings where the input amplifier works at x1 gain. The noise from the signal source itself is an external factor, hence totally variable and mostly unknown. On top of that, the noise of a wideband input amplifier with equalization networks might be anything but white.This is not the end of the theoretical consideration though. In reality, any fast 8-bit ADC, will exhibit an INL (integral nonlinearity) of at least ±0.5 LSB. What if justtheleast significant bit, e.g. the transition from 0 to 1 on the ADC output happens to be off by half a LSB? The answer should be obvious – the amplitude accuracy for signals < -48dBfs would be off by ±6dB.Well, one might think we don’t need high accuracy for levels that low, but I cannot think of many applications where a 1:4 uncertainty could be tolerated.For an FFT analyzer, the ADC does a comparable job as the RF mixer in an SA. And the highest concern for such a mixer is linearity, as this will affect the IPi3 (third order input intermodulation intercept point) and subsequently the intermodulation-free dynamic range. Consequently, the ADC will produce a dense (numerical) fog of distortion and intermodulation products at its output, which the FFT algorithm will show, thus obscuring the real low level signals.Of course, we might be lucky and the particular ADC in our scope might have a relatively low INL for the number range used for close to zero signals, but this is nothing we can rely on. Furthermore, there will always be some nonlinearity, no matter how much some particular ADC exceeds its worst case specifications. So one ADC might produce 10dB lower intermodulation products than some other, but we’ll still never be able to get the intermodulation-free dynamic range as calculated in the textbook.Now for some practical evaluation…

author: Performa01
author_rank: Super Contributor
timestamp: April 17, 2016, 02:36:21 pm
content: 8-bit FFTNow we are going to explore what really can be expected from the FFT in an 8-bit DSO. For all the following examples, full scale is ±2V, which is equivalent to +16dBm with 50 ohms termination at the scope input. All measurements use a ~1.95MHz FFT bandwidth with 2Mpoints, so we have 1M bins each of them being 1.863Hz wide. Blackman Harris window is used throughout and the actual -60dB bandwidth is about 7 times the bin width, resulting in a frequency resolution of some 13Hz.Let’s start with a 990kHz signal at +10dBm without averaging (FFT8_2M_+10dBm)This looks impressive at a first glance. Even without averaging, the noise level appears quite low, but there are lots of spurs and for the reasons already discussed, we cannot know which of them result from the imperfections in the signal itself and which are just due to the shortcomings of the ADC. Even though it would be tempting to relate the more prominent spectral lines to the signal, we just don’t know for sure at levels below -50dBfs.The strongest spur is -48.62dBm and since full scale would be +16dBm it looks like we have a SFDR (spurious free dynamic range) of 64.62dB – some 15dB better than what we’d normally expect from 8 bits.The automatic measurement of the SFDR already hints something different though. The average value reads only 50.76dBc, which would still be 56.76dB total, hence 7dB better than the 8 bit dynamic. But the min/max readings show that the spurious signals change a lot and if we take the minimum of 41.76dBc we get a total SFDR of 47.76dB. But we still don’t know if this SFDR is related to the signal or the measurement system.Now let’s lower the signal level to -10dBm and use averaging to get a clearer picture. The signal related spurs should drop accordingly, whereas the ADC noise should remain largely unaffected (FFT8_2M_-10dBm_avg)The spurs from the signal generator have clearly decreased together with the signal level and are now all the more indistinguishable from the ADC noise. If we look at the SFDR measurement, the uncertainty has decreased to less than 3dB thanks to averaging and the total SFDR is now some impressive 69dB.Now let’s find out the lowest level we can get decently accurate results by lowering the signal level step by step. With a signal level of -30dBm we are at -46dBfs what is close to the limit of the 8 bit dynamic range. No averaging this time, but the screenshot has been taken at a moment where the most prominent spurs were close to a maximum (FFT8_2M_-30dBm_max)The signal level can still be measured quite accurately, whereas the spurious signals have decreased a lot. This could lead to the false impression that they were related to the signal, but in fact they are just part of the intermodulation ‘interference fog’ caused by the ADC nonlinearity. Since every possible ADC output has its own individual INL error, this creates a multitude of distortion and intermodulation products. Now that pretty much only the LSB of the ADC is used, the number of spurs decreases accordingly and even the SFDR reading of the automatic measurement is quite stable at some 12dBc. The total SFDR is therefore some 58.3dB, which is not consistent with the last measurement at a signal level of -10dBm. Let’s investigate that a bit further…What does it look like with a signal level of -40dBm? (FFT8_2M_-40dBm_max)Now there are only a couple of spurs left, but at the same time the signal level has become quite inaccurate. Automatic measurement shows an average of -36.16dBm, which is about 4dB high. Only the minimum reading is close to the true value and there is 6.7dB difference between minimum and maximum.Let’s see if we can get better results by adding some white noise as dither. Due to the INL error of the LSB discussed earlier, a peak to peak noise level of 1 LSB isn’t enough to get accurate readings, hence a much higher level of noise of some 120mVpp had to be applied (FFT8_2M_-40dBm_Noise-18dBm)Now the level display is almost spot-on, but we also start seeing a lot of spurious signals, as was to be expected. After all, we are now able to accurately measure a signal at -56dBfs and the signal is still almost 14dB above noise/spurs, resulting in a total SFDR of nearly 70dB.The dither needs not be white noise, we could try a triangle wave as well. For optimum accuracy, a peak to peak level of 58mV was required and the frequency was chosen at 30Hz (FFT8_2M_-40dBm_Triangle_58mVpp)This throws off the peak level detection of course – the automatic measurement now sees the dither instead of the weak test signal. Other than that, the signal displays correctly and the SFDR is just a little bit worse than it was with the white noise dithering.Now let’s lower the signal level by another 10dB, thus applying -50dBm while still using the 30Hz triangle dither level of 58mVpp (FFT8_2M_-50dBm_Triangle_58mVpp)We still get an fairly accurate level display and there’s only one spurious signal left. So we can actually quite accurately measure a signal at -66dBfs and the signal is still more than 18dB above the spur, resulting in a total SFDR of healthy 84dB. The noise floor is down at <-80dBm, so we could claim a first order dynamic range of more than 96dB – that’s crazy performance from an 8 bit system, is there no difference between textbook theory and practice after all?Well, hopefully it could be demonstrated that the ‘process gain’ – while it has a real effect – isn’t something we can use to make an 8 bit system perform like a higher resolution one. The high number of FFT sample points reduces the bandwidth of the individual bins and lowers the quantization noise, but it cannot prevent the nonlinearity of the ADC from generating lots of unwanted mixing products that immediately rise as soon as the signal level increases and will get particularly bad as soon as the real dynamic range of the 8 bits is entered. Just a look at the screenshots for higher signal levels show that very clearly and the confusing number of spurious signals isn’t exactly my idea of accurate and reliable measurement.Here’s a demonstration of the results for the -50dBm signal without dither (FFT8_2M_-50dBm)The noise level is down to -88dBm and there are only two spurs left, so the picture doesn’t look too bad at first glance, but then the signal level is 6dB low and not stable – the signal variation is more than 8dB.Finally, if averaging is used for the very same situation as above, we get the following picture(FFT8_2M_-50dBm_avg)While the noise level is even lower at <-90dBm and the signals appear much more stable on the screen, the situation hasn’t significantly changed otherwise.

author: Performa01
author_rank: Super Contributor
timestamp: April 17, 2016, 02:44:59 pm
content: 16-bit FFTFrom the previous demonstration, it should be clear by now that an 8-bit system has its tight limitations, no matter what the number of FFT sample points is. While the resolution enhancement can indeed work as long as the right amount of dither is added, which would just be 1 LSB pk-pk if we had an ideal ADC with zero INL, the linearity error of any real ADC will not only require a much higher dither level in order to get accurate results, but also produce a high number of spurious signals that are not random noise, hence cannot be reduced by resolution enhancement and filtering. Only a better ADC (equivalent to a better mixer in a conventional SA) can provide a true improvement in terms of accuracy and usable dynamic range.To prove this, we now look at the FFT of a scope that has a true 16bit ADC. When I say ‘true 16bit ADC’ I actually mean it that way. No other ‘16bit solution’, achieved by some dubious tricks, be it hardware (combining several lower resolution ADCs) let alone software (resolution enhancement by averaging/filtering) can replace the real deal, because none of these tricks will improve the linearity. Particularly in spectrum analysis, linearity is paramount.There is no point in having some virtual ADC that might have 16 bits of resolution, but only 8 bits accuracy/linearity. Just take my word for that.For all the following examples, full scale is ±2V, which is equivalent to +16dBm with 50 ohms termination at the scope input. Many measurements use just 4kpoints, as there’s no need for a ridiculously high number of FFT points just for the sake of noise reduction. Blackman Harris window is used throughout, where the actual -60dB bandwidth is about 7 times the bin width.Let’s start with the same 990kHz signal as before at +10dBm with averaging. This makes sense here, as there are no spurious signals due to ADC nonlinearity to be expected. FFT length is 16kpoints, so we have 8192 bins each of them being 203.5Hz wide. The -60dB resolution bandwidth is therefore some 1.4kHz (FFT16_16k_+10dBm_avg)Now this is finally a measurement that lets us actually judge the signal quality. Even with just 16k points, the noise level is more than 100dB below full scale. The specified SFDR for this scope is >96dB, so we can confidently assume that any spurs down to -80dBm belong solely to the signal. Only below that level, the puzzle whether it’s the signal or the analyzer starts.There aren’t many spurious signals and the strongest one is -59.51dBm, whereas on the 8bit FFT it was -48.62dBm despite the 2M points analysis. But then, it cannot be surprising that we were not able to seriously measure a signal level below 1 LSB of the 8bit ADC back then.The automatic measurements of both the signal level and the (signal related) SFDR are rock solid now, the latter being stable to 0.01dB for 1000 captures!Here’s another example, this time a different signal – 2MHz at +10dBm coming from an analog synthesizer, where we can clearly see the much stronger phase noise compared with the DDS generator used for all the other measurements. This time, Analysis bandwidth is 5MHz and FFT length is only 4k, so we have 2048 bins each of them being 2.441kHz wide. The -60dB resolution bandwidth is therefore some 17kHz. Note that this is also at a different vertical gain, resulting in ±1V and therefore +10dBm full scale (FFT16_4k_+10dBm_avg)Because of the higher sensitivity, we can trust everything down to -86dBm to be signal related. Even with just 4k FFT, we get a detailed picture, low noise level and stable measurements.I stick with the analog synthesizer signal, as this allows me to reach very low levels easily. Now we look at it at a level of -60dBm. FFT length is 4k again (FFT16_4k_-60dBm_avg)With only 4k FFT, the noise level is below -98dBm, so we get a first order dynamic range of 108dB. There are no spurs and the only other signal we see is the 2nd harmonics from the test signal. The automatic measurements are still very stable. Particularly, the signal level is accurate to 0.07dB and has only 0.06dB variation – and that level is only 174µV right now!Well, let’s go down to -90dBm now (FFT16_4k_-90dBm_avg)Now the signal level is -100dB below full scale, so we’re finally leaving the dynamic range of the 16bit converter. We still get an accurate amplitude measurement, even though there is a variation of 3dB now. Why does it work here, when it failed on the 8bit scope? Well, it must have something to do with the dithering, as even in an exceptional low noise scope like this there will be still enough for a 16bit ADC, where the LSB is just about 30.5µV for this particular gain setting.After all, we are measuring a ~7µVrms signal at a full scale of 2Vpp (700mVrms) here, a task where we’d otherwise need at least a 5 digit DMM …Going to the extremes, we now look at -110dBm with 2M points (FFT16_2M_-110dBm_avg)For better visibility, I’ve zoomed to the amplitude range below -93dBm. Noise floor is ridiculously low at about -124dBm now, but at levels that low, the nonlinearities of even a 16bit converter start producing some spurious signals again. Note that there is now a 2nd harmonic only a couple dB lower than the signal itself, that is clearly generated within the ADC.The signal level is still fairly accurately displayed at -110dBm, so a first order dynamic range of up to 120db can be usable under certain conditions. At a signal level of just 2.2µV, automatic measurements are no longer useful, as they now pick mains hum as the strongest signal at a level of ~4.4µV. Well, I don’t know if it is the shielding of the scope frontend or – more likely – just the cabling in my highly mains-contaminated home lab. I forgot to choose a double shielded cable and just used a high quality RG58 for my tests.

author: Performa01
author_rank: Super Contributor
timestamp: April 17, 2016, 02:51:02 pm
content: 2 Tone ComparisonUp to now, we’ve just examined the first order dynamic range and the test scenarios didn’t really reflect any practical application. No one would look at a single signal far below full scale of the instrument, rather increase the sensitivity to make good use of the dynamic range available. In case of the first order dynamic range this is actually full scale or a couple dB below the 1dB compression point on a real SA.Much more relevant is the analysis of small signals in presence of a large one. This applies to measurements of signal harmonics (of course sub- and non-harmonics as well), modulation spectra and phase noise. So if there’s always at least one strong signal, we might get away without an additional dither for the 8-bit system maybe?Let’s start with 8 bits and two signals at 990kHz, +10dBm and 1010kHz, -40dBm. Full scale is +16dBm and 2M points are used again. FFT bandwidth is 1.953MHz. The required dynamic range is 56dB, thus exceeds what would normally be possible with just 8 bits (FFT8_2M_+10dBm_-40dBm)Well, this doesn’t look too bad and is actually quite usable. For some unknown reason, automatic measurement of the peak level has a strong variation which isn’t real – I probably forgot to reset the statistics after setting up the signals.Now decrease the level of the 2nd signal to -50dBm (FFT8_2M_+10dBm_-50dBm)The result is still usable, so we have just proved that we can work within a 66dB first order dynamic range. However this result cannot be generalized, as it heavily depends on the individual properties of the ADC. This particular one seems to be quite good in terms of INL near zero, but there’s no guarantee that all 8bit scopes will behave the same in this regard – not even all scopes of the same type from the same manufacturer.Now let’s take it to the extremes and lower the 2nd signal level once more to -60dBm (FFT8_2M_+10dBm_-60dBm)Well, the good news is that the measurement of the 2nd signal level is still accurate, but it now submerges in the jungle of spurious signals. Not a problem as long we know exactly what signals to measure, but we won’t be able to identify unexpected spectral components like non-harmonics when analyzing a signal. So as much as I would love to conclude that we can get a usable dynamic range of 76dB from just 8 bits, it is not really true. While it basically works, it is no fun for sure to hunt for your wanted signal in the noise, even though that particular noise has a discrete spectrum that leaves slots for the test signal to fit into.In comparison, the 16bit FFT has no troubles with a signal level of -70dBm. At an FFT length of 2M points we need no averaging to get a noise floor close to -100dBm and we can be confident that all spurs down to at least -80dBm are real, i.e. belonging to the signal itself. In the following screenshot, we can even see the phase noise of the 990kHz signal quite clearly (FFT16_2M_+10dBm_-70dBm)Now the question might arise, how does that compare to a real SA? Well, as much I’d love to demonstrate that on a R&S FSEA30, right now I only have a cheap USB spectrum analyzer (Signal Hound SA44). While amazingly useful, it still has lots of limitations but nevertheless is good for a very descriptive demonstration.So here it goes. One of the limitations is that the span must not exceed 1MHz so to get at least 200Hz RBW and this still takes a sweep time of nearly 4 seconds. Nothing like 210ms for a 5MHz sweep with 30Hz RBW as I did on the 16bit scope…So I used a slightly different setting for various reasons. The signals are now at 990kHz and 1.5MHz and the levels are +10dBm and -80dBm, a 90dB difference! (SA_RBW200Hz_+10dBm_-80dBm_avg8)Now this is very revealing. We can clearly see that a traditional superheterodyne receiver with an IF bandwidth of 500kHz is ‘swept’ (probably more like ‘stepped’) and an FFT processing is done to provide the narrower resolution bandwidths. The dynamic range of the superhet is quite good and clearly exceeds 100dB at 200Hz RBW. But the FFT shows a signal dependant noise floor, which makes the signal source look much worse than it is. From the +10dB 16bit FFT of the DDS signal we know that the phase noise doesn’t start any higher than -90dBm and looks completely different to what we see here. One of the reasons for the high noise floor might be the flat top window, which has less side lobe suppression compared to Blackman Harris. Apart from that, I strongly suspect that the ADC might only be 14 bits.Of course, if we lower both the span and the RBW, FFT noise will drop too, but for wider spans, this SA clearly cannot compete with the 16bit FFT.With a Span of 50kHz and a RBW of 25Hz (13Hz would be possible too), we can resemble the original setup, i.e. make the 2nd signal 1010kHz again. The noise floor is now low enough to pass the test with -80dBm for the 2nd signal (SA_RBW25Hz_+10dBm_-80dBm_avg8)The two-tone intermodulation tests can be found here:https://www.eevblog.com/forum/testgear/fft-spectrum-analysis-reviewed/msg921837/#msg921837

author: uncle_bob
author_rank: Unknown
timestamp: April 17, 2016, 03:02:59 pm
content: HiNicely done. There really is no "free lunch" in this case.There are other studies comparing 16 bit ADC based systems to (older) "Real SA's". In most cases, the modern 16 bit ADC wins out. Of course, that only works up to a limited frequency point due to sample rates. Digitizing a 22 GHz spectrum at 44 Gs/s and 16 bit samples is not quite possible (yet).Bob

author: Performa01
author_rank: Super Contributor
timestamp: April 17, 2016, 03:32:49 pm
content: Hi Bob,thank you!I am not at all surprised about that studies. After all, most 'real' SAs only have a SFDR of some 70dB. That's also why I particularly mentioned the R&S FSEA 30 - which I will get eventually, once I've rebuilt my home lab to get a larger bench and deeper shelves, that isThe real deal would be the professional 24 bit audio interfaces, but then their frequency coverage is just too limited for anything other than audio...

author: Kleinstein
author_rank: Super Contributor
timestamp: April 17, 2016, 03:50:31 pm
content: There are spectrum analyzers that combine classical heterodyne and FFT: so the heterodyne is used to bring the signal down to a range where the higher resolution ADC works well and than use FFT to get the high resolution / small RBW and fast "scan". To get the real dynamics and less spurious they need the higher resolution ADCs, though they might get rid of some of the spurious signals using more than one IFs.

author: Performa01
author_rank: Super Contributor
timestamp: April 17, 2016, 04:08:38 pm
content: Quote from: Kleinstein on April 17, 2016, 03:50:31 pmWell, that's exactly what the SA44 does. But as I wrote, I'm not entirely sure if that device actually utilizes a 16 bit ADC - it might be just 14 bits. Nevertheless, the performance isn't bad at all, especially considering its pricepoint.The third order dynamic range is only about 65db, but then again, many of the old boat anchors haven't been any better either - and this doesn't even seem to have changed a lot...

author: uncle_bob
author_rank: Unknown
timestamp: April 17, 2016, 05:16:23 pm
content: Quote from: Performa01 on April 17, 2016, 04:08:38 pmHiThe issue is still that analog mixers (plus amps plus filters) are limited in their IP3 performance. That has not changed a whole lot over the last decade or two. A relatively cheap modern 16 bit ADC does better on IP3 than the mixer chain you can afford to put in a piece of test gear. That assumes the test gear costs less a pretty nice new car. Of course if the sky is the limit, there are some pretty crazy ADC's out there ....Bob

author: hendorog
author_rank: Super Contributor
timestamp: April 17, 2016, 09:08:36 pm
content: Quote from: Performa01 on April 17, 2016, 04:08:38 pmFWIW, the new software Spike doesn't support anything less than 6.5kHz RBW below 16MHz on my SA124B.I don't know why that is, but it made me think that perhaps you are in a frequency/span where the SA is particularly weak?Also I get much better sweep rates with Spike, and just about everything else works much better so its well worth the upgrade - if you can get off XP.

author: egonotto
author_rank: Super Contributor
timestamp: April 18, 2016, 03:23:08 am
content: HiQuote from: Performa01 on April 17, 2016, 02:44:59 pmPlease can you say which PicoScope oscilloscope you use.Do you think that the PicoScope 5243A has a true 12 Bit till 16 Bit ADC.Thanksegonotto

author: ADT123
author_rank: Regular Contributor
timestamp: April 18, 2016, 07:26:48 am
content: Quote from: egonotto on April 18, 2016, 03:23:08 amI just did some captures with a couple of PicoScopes and also a LeCroy WaveAce 102 in all cases the input was a 1.8V pk-pk 20kHz low distortion sine wave. SFDR figures:LeCroy WaveAce 102 50dB (approx)PicoScope 5244B 68dB (8 bit mode)PicoScope 5244B 78dB (16 bit mode)PicoScope 4262 93dB (16 bit)One thing I found was that at 16 bits the quality of the BNC cable matters. Swapping a cheap (foil wrapped internally) cable for a nice braided one gave a couple of dB extra. Suspect the 5244B would have got 80dB.The PicoScope 5244B does change its resolution in hardware rather than software trickery (seehttps://www.picotech.com/library/oscilloscopes/flexible-resolution-oscilloscope) but does not get as good a result as the 4262. I suspect this is because the PicoScope 5000 is a 1GS/s, 200MHz scope whilst the PicoScope 4262 is a low frequency 5MHz scope optimized for high dynamic range.Will post screenshots in a monent

author: ADT123
author_rank: Regular Contributor
timestamp: April 18, 2016, 07:41:01 am
content: Screenshots from the last post. One tip that can help when using scopes that offer large numbers of FFT points is to use the maximum number of points and a higher bandwidth than you want to view. Next zoom in on the area of interest. In these examples the signals I want to see are below 1MHz but by doing an FFT to 5MHz the higher frequency noise is not folded back into the area of interest so lowering the noise floor.Lecroy - about 50dBPicoScope 5244B in 8 bit mode. 68dBPicoScope 5244B in 16 bit mode. 78dBPicoScope 4262 16 bit. 93dB

author: Performa01
author_rank: Super Contributor
timestamp: April 18, 2016, 04:29:08 pm
content: Quote from: hendorog on April 17, 2016, 09:08:36 pmWell, I don't think the SA124 can be compared to the SA44, as the latter explicitly goes down to 1Hz. Also, my experience with this neat little device did not hint on any particular weakness in the lower frequency regions, quite the contrary. If anything, noise floor gets notably worse above 3GHz.A RBW of 6.5kHz sounds really odd, particularly on the lower frequencies. Maybe the high bandwidth SA124 actually has some limitations at its lower bandwidth limit...And yes, for various reasons that would be off topic to explain here, I'll stick with Win XP on my lab computer, so the latest Software isn't an option for me. Maybe I'll take my company notebook to my lab some day, to check how much of an improvement the new Spike software actually is for the SA44...

author: Performa01
author_rank: Super Contributor
timestamp: April 18, 2016, 04:51:57 pm
content: Quote from: egonotto on April 18, 2016, 03:23:08 amHi,it is a 4262, since this is the only true 16bit realtime scope I'm aware of.The 5000-series 'variable resolution' scopes are certainly an improvement, but it involves combining several lower resolution ADC cores. While this gives much better results than any 8-bit DSO, it is still not as good as a true 16 bit ADC. Just look at the dynamic performance specifications published by Pico Technology:ScopecrosstalkTHDSFDRNoise426250000:1-95dB96dB8.5µVrms5000400:1-70dB70dB70µVrmsThe maximum bandwidth is limited to some 31MHz for the 5000 series in 16bit mode, whereas it is just 5MHz on the 4262. But all the other specs are much better on the 4262. For me, it is just the right tool for low frequency signal analysis at a very attractive price.

author: bozidarms
author_rank: Regular Contributor
timestamp: April 18, 2016, 07:13:06 pm
content: Rigol DS4012(4052) ------------ FFT about 70dB (without trace averaging)

author: hendorog
author_rank: Super Contributor
timestamp: April 18, 2016, 07:39:48 pm
content: Quote from: Performa01 on April 18, 2016, 04:29:08 pmYes I understand being stuck on XP - had the same problem but eventually bit the bullet. Now I'm stuck on Windows 7...The SA44 also has this limitation - this is from the SA44 manual:"For spans larger than 99MHzor sweeps that start below 16MHz, the RBW can be set between 6.5 kHz and 250kHz""All RBWs are available for spans 200kHz or less."My thought was that the old software may not be indicating this limitation correctly.If you are lucky the limitation may not be present in the older software, but that would be odd - why add a new limitation?

author: G0HZU
author_rank: Super Contributor
timestamp: April 18, 2016, 08:08:15 pm
content: QuoteI have several decent (but ~30yr old) RF spectrum analysers here covering up to 22GHz but I still use the FFT feature on my little Tek TDS2012 from time to time for certain tasks. However, since I bought an Agilent E4406A vector signal analyser I tend to use the FFT feature on the TDS2012 less.The TDS2012 can do some things that the big old spectrum analysers can't such as look at very narrow bandwidth RF signals that require sub 10Hz or sub 1 Hz RBW.A typical test signal would be an on/off keyed signal at 70MHz that is being keyed at a 1Hz rate. The TDS2012 can display the spectrum of this with RBW <1Hz and so can the E4406A but none of the other analysers can do it 'properly'.A typical ham/CB use for the FFT feature in the TDS2012 would be to look at narrowband AM/FM/SSB signals up to 145MHz using either test tones or human speech. The displayed range is only 50-60dB but this is usually enough to see if a transmitter is basically OK or if it has an issue with excessive bandwidth. It can refresh the display quickly in this mode and show speech peaks etc. Not quite real time but close enough and MUCH quicker than a typical swept spectrum analyser.

author: hendorog
author_rank: Super Contributor
timestamp: April 18, 2016, 08:34:14 pm
content: Quote from: bozidarms on April 18, 2016, 07:13:06 pmWith trace averaging set to max (8192) noise floor is down below -100dB

author: G0HZU
author_rank: Super Contributor
timestamp: April 18, 2016, 09:50:21 pm
content: QuoteI've not used a Signalhound analyser but I think that at least some of the noise on the Signalhound plots may be the noise profile of the PLL used for its local oscillator. Try measuring the Signalhound up at 3 or 4GHz with a similar span and this noise level will be much higher because the 1st LO will be much noisier up at these frequencies. I'd expect it to be in the ballpark of 30dB higher. But that's just a guess based on how I think the Signalhound 44 LO system is designed

author: Performa01
author_rank: Super Contributor
timestamp: April 19, 2016, 01:16:42 am
content: Intermodulation TestsWe still have just examined the first order dynamic range so far. But there are also applications, where we want to measure spectral components with tiny amplitude in presence of more than just one strong signal, e.g. when observing modulation spectra but also – and primarily – when testing mixers, amplifiers, attenuators and the like for their intermodulation performance. After all we want to measure the performance of the device under test – and not of the analyzer. This is where the third order dynamic range becomes important, and this will always be worse than the first order one.We start with the 8 bit FFT, feeding two tones at 990kHz and 1010kHz and a level of -13dBm. The scope is in the ±200mV full scale range, which is equivalent to -4dBm. FFT length is 2M points and FFT bandwidth has been chosen to ~4.5MHz. The picture is zoomed so we can clearly see what’s going on near the test signals (FFT8_2M_Two_-13dBm_avg_zoom)Once again, this looks surprisingly good. Both 3rd order intermodulation products at 970kHz and 1030kHz are at -85dBm and it looks like we have no troubles measuring them, even though they are at -81dBfs, hence far outside the 8bit dynamic range.Looking at the whole picture, we can see that there are indeed stronger spurs at odd frequencies, not related to the test signals or their harmonics and associated intermodulation products (FFT8_2M_Two_-13dBm_avg)Nevertheless, this test provided a 3rd order dynamic range of 72dB, thus competing with many ‘real’ spectrum analyzers.Now for the 16bit FFT. Same settings as before, except for the FFT bandwidth now being exactly 5MHz. Here is the zoomed picture again (FFT16_2M_Two_-13dBm_avg_zoom)There is little guessing now. The intermodulation product at 970kHz is much stronger at -107.7dBm and also slightly outside the genuine dynamic range of the ADC. There are only few spurs, some of them a little stronger at -105dBm. The whole picture reveals nothing new, just the harmonics of the test signals together with their associated intermodulation products. All in all we get an intermodulation-free dynamic range of 94.6dB and only very few ‘real’ spectrum analyzers can beat that.Should anyone wonder what this two-tone test signal looks like in the time domain, here it is (FFT16_2M_Two_-13dBm_time)Pretty much a 100% amplitude modulated waveform. A real (analog) amplitude modulator has to be very good and tweaked a lot in order to produce a DSB (double sideband) signal like this – in fact it is next to impossible. A little bit of the carrier will always be left. But the reverse direction works beautifully. Just mix the two sidebands together and there is no delicate carrier suppression anymore we have to worry about, bingo!Finally, let’s have a look at the ‘real’ SA. These devices are much more sensitive than any DSO, but conversely can’t handle high signal levels. Even though -13dBm is not a problem for the SA as far as the first order dynamic range is concerned, however this level is too high for the 3rd order dynamic range test. Usually, -20dBm is used to determine the third order intermodulation intercept point and that’s exactly what we want to do now.Very similar setup as before, two tones at 990kHz and 1010kHz, both at -20dBm, but due to the restrictions of the SA the scan is limited to the range from 800kHz to 1.2MHz and RBW (resolution bandwidth) has to be much higher at 100Hz (SA_RBW100Hz_Span400kHz_Two_-20dBm_avg16)Now this is gorgeous. We get a 3rd order dynamic range of 91dB right from the start!This would be equivalent to an IPi3 of +20dBm, which is 10dB higher than e.g. on a Rigol DSA 815.But this is not the entire truth. This looks like the FFT results we’ve seen before, so we have to repeat the test with the two tones further apart in order to force the SA44 to do a real scan. So the tones are now at 1.2MHz and 1.7MHz and the span has been increased to 2MHz. By this, the RBW has increased to 400Hz as well (SA_RBW400Hz_Span2MHz_Two_-20dBm_avg16 )Now we’re talking! The 3rd order dynamic range is only 60dB now, but the intermodulation products are still well above the noise floor, so that is not the end yet. Let’s calculate the IPi3 first – it is the familiar +10dBm! As the noise floor is about -105dBm, we could lower the signal amplitudes by 10dB and the intermodulation products would submerge in the noise at -110dBm. But we could just use the internal attenuator instead (SA_RBW400Hz_Span2MHz_Two_-30dBm_avg16)Bingo! The signal levels appear unchanged, but interestingly the noise floor has dropped too despite activating a 10dB internal attenuation. i.e. the NF (noise figure) seems to have even improved – quite the opposite to what I would have expected. My only explanation is that the displayed noise isn’t real noise from the frontend, but just noise from the ADC. If anyone has a better explanation, I would like to hear (read) it.Either way, the intermodulation products have dropped by 20dB as expected and thanks to the fact, that the noise floor has not increased, the third order dynamic range is now respectable 80dB.With the internal 10dB attenuator, the IPi3 is +20dBm of course, which wouldn’t mean a thing were it not for the not elevated noise level, which actually gives us a rather decent intermodulation-free dynamic range.So in the end I have to apologize to the SA44, which I have previously accused for only having 65dB 3rd order dynamic range. In fact, it is much better than that – at least at lower frequencies.

author: Performa01
author_rank: Super Contributor
timestamp: April 19, 2016, 01:39:28 am
content: Quote from: ADT123 on April 18, 2016, 07:41:01 amThat's some excellent suggestion!That's also a reason why it's nice to be able to use a high number of sample points for the FFT, so we can avoid aliasing and still preserve a low RBW.QuoteWell, your signal source appears to be very clean indeed - but are you really sure it isthatgood? You are measuring the 3rd harmonic of the test signal, which in theory could be generated by the ADC nonlinearity, but this is unlikely, since it is still within the genuine dynamic range of the 16-bit ADC and Pico Technologies specify <-95dB THD for this scope. So I would rather suspect it's coming from the signal source...EDIT: Btw, I've just noticed, you have the internal signal generator on. I've found that this also produces some spurs, even when the siggen output is not connected. So for high sensitivity measurements it should be turned off.

author: Performa01
author_rank: Super Contributor
timestamp: April 19, 2016, 02:03:26 am
content: Quote from: G0HZU on April 18, 2016, 09:50:21 pmWell, here's one for 1GHz (SA44_RBW200Hz_Span1MHz_1GHz_-10dBm_avg8)Right at the moment I do not have any decent signal generator for 3GHz, so I have to stick with something of unknown qualities. But if the signal happens to be ok, then your guess would be about right (SA44_RBW200Hz_Span1MHz_3GHz_-10dBm_avg8)

author: egonotto
author_rank: Super Contributor
timestamp: April 19, 2016, 02:23:24 am
content: Hi,thanks for the answers.Regardsegonotto

author: hendorog
author_rank: Super Contributor
timestamp: April 21, 2016, 12:55:38 am
content: So to determine the source of the noise in the last post, is it simply a case of adding external attenuators and comparing?i.e. If the level of noise doesn't drop then it's coming from the SA. If the level of the noise does drop by the same amount as the attenuation then it's coming from the signal source.To test that theory, here is the SH tracking gen connected to my SA124B first with no external attenuators, and then with 40dB of external attenuators.The plan is to determine if the noise is coming from the tracking generator or the spectrum analyser.First image is with no external attenuators and reference set to 0dBm. It shows noise close to the signal starting at around -60dBm.Second image is with added 40dB external attenuators, and reference decreased by 40dB. This shows noise close to the signal at around -100dBmTherefore the noise is coming from the TG and not the SA - is this correct?Third image shows just the noise floor - I just moved the TG off frequency to 3.5 GHz.

author: Performa01
author_rank: Super Contributor
timestamp: April 21, 2016, 06:36:23 am
content: Quote from: hendorog on April 21, 2016, 12:55:38 amNo, quite – your test doesn’t tell the whole story.If you lower the signal level by means of external attenuation, the SNR of the signal source remains unchanged. Now you decrease the reference level of the SA which means nothing else than increasing the IF gain by the same amount, thus pronouncing the SA mixer noise.As we see almost the exact same picture as before, that means the noise from the SA frontend is negligibleThe point is, when evaluating the noise at 3GHz, we were suspecting the LO and not the mixer to be the noise source. The LO signal is getting multiplied by the input signal, hence LO phase noise will not be visible with no input signal present. Consequently, you get a fairly clean noise floor with no input.That also means the two test scenarios (with and without attenuation) don't change anything in terms of LO noise, because its visibility is closely related to the IF level, which remains the same.

author: Performa01
author_rank: Super Contributor
timestamp: April 21, 2016, 06:41:42 am
content: Without a proper 3GHz signal source with known (good) qualities, it’s hard to evaluate the SA44 phase noise performance at that frequency.The only idea to get a little closer to the answer is to analyze the harmonics of a 1GHz signal. First the fundamental at 1GHz (SA44_RBW200Hz_REF-10dBm_1GHz_-20dBm_H1_avg8)Ignoring the weak sidebands, the phase noise is almost down -80dBc and we would expect to be the harmonics of similar quality, just the frequency spans multiplied by the order of the harmonic.We can see the 3rd harmonic at some -75dBm and noise floor is <-110dBm. That is pretty quiet even at 3GHz, despite the mixer still gets the -20dBm 1GHz signal, which once again confirms that the mixer is working very well. But we still don’t know about the LO.The earlier tests show pretty strong phase noise up to some -48dBc, and this would still be hidden in the picture above. So let’s just increase the signal level by 20dB to 0dBm, thus supposedly being near the compression point of the mixer (SA44_RBW200Hz_REF-10dBm_1GHz_0dBm_H3_avg8)Now that is revealing! The 3rd harmonic is now -40dBm, so it has increased by 35dB, that is 15 dB more than the fundamental, clearly hinting that we have indeed hit or exceeded the CP, thus slightly overloading the mixer and let it generate some harmonics by itself.But then, it doesn’t really matter where the signal comes from. What we see on the display is the IF, which includes the LO signal whose level hasn’t changed, but is now more pronounced as the mixer product has increased.And we can see indeed the phase noise starting to emerge. Now lowering the reference level to -20dBm should give an even more clear picture (SA44_RBW200Hz_REF-20dBm_1GHz_0dBm_H3_avg8)Phase noise starts at about -70dBc, so it is much better than what we got from the earlier tests – and it is only 10dB worse than at the 1GHz fundamental. So it’s not too bad…

author: uncle_bob
author_rank: Unknown
timestamp: April 21, 2016, 11:27:58 am
content: HiOne of the most confusing things about looking at phase noise on a spectrum analyzer is bandwidth normalization. Phase noise has a "nose density" in dbc / Hz. Change the bandwidth and you change the dbc. In some cases *knowing* the actual noise bandwidth of the filter used is the first challenge. You then get into some cute conversion issues in addition to the basic "10 * log(square root of the bandwidth)" calculation.The net result is that I run a test on my analyzer and get -90 dbc. You run a test on your analyzer and get "only" -80 dbc. Bob obviously wins the race and you buy the beerWell, not so much. If Bob fiddled the bandwidth so his is narrower, he just cheated you out of a beer ...Bob(Thanks for the beer)

author: ADT123
author_rank: Regular Contributor
timestamp: April 21, 2016, 12:13:08 pm
content: Quote from: Performa01 on April 19, 2016, 01:39:28 amI tried using the internal signal generator on the PicoScope 4262 and also a Blackstar low distortion audio signal generator. Both gave very similar results and to be honest I can not remember which I used!It is not easy making such sensitive measurements - I dont know if the signal source or scope is the limit here. Good tip about making sure its turned off when not in use. As mentioned even the quality of the BNC cable matters when you are measuring these levels. Suspect if a cat farts in the next town it will get detected!

author: Performa01
author_rank: Super Contributor
timestamp: April 21, 2016, 01:05:51 pm
content: Quote from: uncle_bob on April 21, 2016, 11:27:58 amI’d gladly send you some good Austrian beer – even though you have cheated – but I’m afraid it would be pretty warm by the time it arrives at your place…Well, the original topic of this thread was FFT, now we’ve finally arrived at phase noise measurement on various signal sources. It shouldn’t be too difficult if the phase noise specifications of the equipment involved is known. Sadly the specs of the SA44 are rather incomplete, to say the least.But my signal generator is well specified - and I was able to measure it quite nicely with the 16bit scope, so let’s evaluate that. The 2nd screenshot in my 16bit FFT post shows that signal generator at 2MHz and we can see some -85dBc of phase noise for frequencies up to 150kHz from the carrier, at a resolution bandwidth of ~17kHz. So the phase noise reads some 42dB worse than with 1Hz RBW and we get a total phase noise of -127dBc/Hz.And what does the specification say? It states -126dBc/Hz at any distance 20-500kHz from the carrier. Bingo! We now know (and have verified) the performance of the signal source.The neat thing is, according to the specifications, this generator maintains the phase noise over its entire bandwidth (within a few dB tolerance), so we can expect a similar phase noise at 1GHz.But we can stay at 2MHz anyway, as the first screenshot with the SA44 shows. At only 200Hz RBW we get a phase noise of some -85dBc again. Because of the 200Hz RBW we can add 23dB and end up with some -108dBc/Hz, which is 18dB worse than the generator signal alone.But then, it’s not so uncommon that low end SAs (even from reputable brands) have such a high phase noise that they are pretty much useless for phase noise measurements.

author: G0HZU
author_rank: Super Contributor
timestamp: April 21, 2016, 03:38:16 pm
content: The other option is to simply make a basic test oscillator at 4.4GHz to test the phase noise. The noise performance at 100kHz offset would only have to be 'fairly' decent to be better than the LO phase noise on all but the best spectrum analysers.eg you should easily be able to make something that could deliver -120dBc/Hz phase noise at 100kHz offset at 4.4GHz using a basic free running oscillator with a printed resonator. Maybe half a dozen parts on the PCB in total?

author: Ivan7enych
author_rank: Regular Contributor
timestamp: April 21, 2016, 05:49:33 pm
content: Quote from: G0HZU on April 21, 2016, 03:38:16 pmCan you tell a bit more how to make this basic test oscillator ?

author: G0HZU
author_rank: Super Contributor
timestamp: April 21, 2016, 06:56:34 pm
content: You can throw some basic physics at it and design for a resonator power of 10dBm and a device noise figure of 6dB and a loaded Q of 20. At 100kHz offset you should see about -120dBc/Hz phase noise if you put these numbers into a basic version of Leeson's equation.In reality, the results may vary by about 5dB and a lot depends on if you can control/set the zero phase point around the loop at 4.4GHz to coincide with the peak in group delay in the resonator. Get this wrong and the phase noise will degrade a lot.But the circuit would just be a cheap MMIC (50 ohm in/out) amplifier from MiniCircuits on a tiny piece of decent PCB material (Rogers 4003 or 4350) with a simple printed resonator shape between the input and output of the MMIC amplifier and a couple of coupling caps tapped into the resonator and also a SMD choke and resistor network to provide the supply voltage. The art is in getting the phasing correct and in preserving the loaded Q. The best way to do this is to model the circuit on a PC.But the noise performance of a basic oscillator like this is quite easy to predict. The phase noise at 10kHz offset will be about 20dB higher than at 100kHz but there will be some additional flicker noise from the active device. So you might see -97dBc/Hz at 10kHz offset and -120dBc/Hz at 100kHz offset from such an oscillator at 4.4GHz. But you would need a very good spectrum analyser to be able to measure the noise performance at 100kHz offset.

author: hendorog
author_rank: Super Contributor
timestamp: April 21, 2016, 10:20:46 pm
content: In the spirit of uncle_bob, lets cheat:I have one of these MLSL-0406IC which I got from eBay a couple of years ago.http://www.admiral-microwaves.co.uk/pdf/micro-lambda/MLSL_IC_SeriesSheet_2GHz.pdfThe specs seem to match up with those in G0HZU's post.10kHz: -98 dBc/Hz vs -97100kHz: -122 dBc/Hz vs -120Here are some plots at about 6GHz from it - (I don't have anything setup to change the frequency of it right now so its stuck at ~6Ghz)So I guess either the SH cannot display the PN that the synthesizer is capable of, or the synth isn't working to spec?

author: uncle_bob
author_rank: Unknown
timestamp: April 21, 2016, 11:39:12 pm
content: HiOk, so if spectrum analyzers sometimes do and sometimes do not show you the real phase noise .... how do you settle the bet? After all, there *is* beer being drunk and *somebody* must pay for it allOne fairly easy way is to do a quadrature phase noise setup. You take two identical oscillators and run them into a mixer. Lock them up so the DC output of the mixer is zero volts. Net result... that messy carrier is now gone (actually it's DC, but that's not part of the bet). The noise on the oscillators has been folded together so you have both sidebands of both oscillators coming out the IF port of the mixer.Next step is to calibrate the gizmo. It's a phase detector in this case, break lock and you have a nice predictable sine wave with 360 degrees phase coming out. Work out the degrees per volt around zero volts with a scope. You have your calibration factor.Lock it back up and take a look at the output with a *low frequency* spectrum analyzer. You no longer need super duper dynamic range. You just need a really good noise floor.Now you can prove what the noise is and get that slacker Bob guy to buy all the beer ...Bob(quietly leaving the bar by a side door ... burp ...)

author: tautech
author_rank: Super Contributor
timestamp: April 22, 2016, 08:46:31 am
content: Just love threads like this......thanks for sharing the knowledge guys.Subscribed.

author: G0HZU
author_rank: Super Contributor
timestamp: April 22, 2016, 12:18:29 pm
content: At work I'm lucky to have access to some nice RF test gear and I grabbed one of the E5052A SSAs today and measured a homebrew version of the 4.4GHz oscillator outlined above.http://www.keysight.com/en/pd-409739-pn-E5052A/signal-source-analyzer-10-mhz-to-7-265-or-110-ghz?cc=US&lc=engI used a MiniCircuits GALI -49 as the amplifier and the PCB was built on a tiny bit of Rogers 4003C 0.02" thick. It's not quite on 4.4GHz so would have to be trimmed down a bit to get inside the 4.4GHz limit of the Signalhound analyser but see below for a phase noise plot of this oscillator.You can see that it agrees very well with theory! However, I did have to bias the GALI 49 at 80mA to get the phase noise this good. It was at -117dBc/Hz at 100kHz offset with 50mA bias. There are often hidden factors like resonator losses that degrade the phase noise a bit. So usually the result comes out slightly worse than the prediction using a very simple version of Leeson's equation. Close enough though!I can post up pics of the PCB and the initial design simulation if anyone wants to see them?

author: Performa01
author_rank: Super Contributor
timestamp: April 23, 2016, 10:42:41 am
content: Uncle Bob (the one who likes beer so much) mentioned noise bandwidth of the filters – and now I have to admit that I was a bit sloppy about that when evaluating the phase noise of my signal generator. Back then, I just took the -60dB bandwidth, which clearly is much wider than the actual noise bandwidth.So how do we know what the noise bandwidth in our FFT analysis is?FFT can be viewed as being equivalent to a number of bandpass filters, that is half the number of sample points. These filters obviously are pretty steep and from the textbook we know that in this case the noise bandwidth will roughly be the same as the -3dB bandwidth.So what’s the actual -3dB bandwidth in our FFT then?This depends on the window function we’ve chosen. Pico Technology was kind enough to show a table of all available window functions in their user manual, so I’ll repeat the relevant information here:WindowRel. BWSide LobeBlackman1.68-58dBGaussian1.33 ~ 1.79-42 ~ -69dBTriangular (Barlett, Fejer)1.28-27dBHamming1.30-41.9dBvon Hann1.20 ~ 1.86-23 ~ -47dBBlackman-Harris1.90-92dBFlat-top2.94-44dBRectangular0.89-13.2dBAs can be seen from the table above, Blackman-Harris is the general purpose window, particularly if a high dynamic range is required, where it is the only window function with a reasonable side lobe suppression (apart from the Kaiser-Bessel window with high values for the Bessel parameter , which is not available in the Pico software).Most of these window functions have their specific uses because of specific merits, but due to their poor side lobe suppression cannot be recommended for any applications where dynamic range is of importance. This is particularly true for rectangular and triangular, as they create a lot of noise even in an 8 bit system. Rectangular has the narrowest bandwidth and maximal sharpness of all window functions though.The performance of Gaussian and von Hann windows depends on window-specific parameters, which aren’t specified in the Pico documentation. This is most likely true with other scope manufacturers as well, where we can select these window functions from a menu but don’t actually know what we get. The table above just lists the performance for the usual range of these parameters. Consequently, these window functions should be avoided unless someone actually evaluates their specific properties on the scope in question beforehand.It is clear, we should use Blackman-Harris for the phase noise measurement, but first let’s check whether the -3dB bandwidth of 1.9 times the bin width as specified in the table is plausible. The following screenshot shows both the -3dB and -60dB bandwidth by cursor measurement on the high frequency side of the filter curve. This is the zoomed picture from a 64k FFT at 5MHz bandwidth and 152.6Hz bin-width (FFT16_64k_2MHz_+3dBm_avg_zoom)The automatic measurements confirm that the signal is indeed 2MHz at +3dBm. The -3dB point is at 2.000152MHz, hinting on a total -3dB bandwidth of 2 x 152Hz = 304Hz. Using the factor 1.9 from the table, we get 152.6 x 1.9 = 290Hz. Well, close enough. The difference is most likely due to a slight asymmetry in the top of the filter curve.The -60dB point is at 2.0005631MHz, hence -60dB bandwidth is 2 x 563Hz = 1126Hz. That’s pretty steep indeed, but is still some 3.75 times the noise bandwidth, thus introducing almost 6dB error if this is used for noise calculation.I will use 300Hz as the noise bandwidth for the following measurement.Here’s the 64k point 16 bit FFT of a 2MHz signal from the analog synthesizer at +4dBm, slightly zoomed to show the range 1MHz to 3MHz (FFT16_64k_2MHz_+4dBm_avg)I’m measuring the point of strongest phase noise here, which is at about 180kHz distance from the carrier. At only 20kHz distance, it would be some 4dB better.Worst case phase noise at that frequency is -96.2dBc for 300Hz bandwidth. Normalized to 1Hz we can add 10 log(300) = 24.8dB, so we get a phase noise of -121dBc/Hz.The specification of the synthesizer says -120 to -126dBc for the entire frequency range and the phase noise curve is only given for 480MHz, where it is indeed a flat line at -125dBc from 20kHz to some 500kHz distance from the carrier. So at least at the lower end of the frequency range, phase noise is 4dB higher than that, which is still low – certainly lower than what most ‘real’ spectrum analyzers can measure.

author: G0HZU
author_rank: Super Contributor
timestamp: April 23, 2016, 11:32:00 am
content: QuoteAt a wanted carrier frequency of 2MHz that level of noise performance is very poor for a signal generator. There are much better generators available than that (for 2MHz carrier frequency) or you could make something using junk box parts.I'd expect a fairly basic/cheap homemade 2MHz LC oscillator to achieve:>-130dBc/Hz at 1kHz offset>-155dBc/Hz at 10kHz offset>-175dBc/Hz at 100kHz offsetOr you could try testing a 10MHz crystal oscillator divided by 5 to get 2MHz. This should give better than -150dBc/Hz at 10-100kHz offsets. eg use a 74HC390 to do the divide by 5.QuoteDown at these test frequencies of a few MHz you can use a regular lab (microwave) spectrum analyser and a crystal filter to allow simple phase noise measurements down at -140dBc/Hz at just a few kHz offset.Again, it is fairly simple to design/build a 2MHz crystal filter that is a few kHz wide

author: G0HZU
author_rank: Super Contributor
timestamp: April 23, 2016, 11:55:16 am
content: When I first bought my E4406A analyser I experimented with direct injection at the 14 bit ADC stage and was able to get -136dBc/Hz phase noise at just a few kHz from the carrier.https://www.eevblog.com/forum/testgear/agilent-e4406a-vector-signal-analyser/See the image below.This is set at 15dB/div and not 10dB/div. Just to clarify, I have bypassed the RF converter stage in the E4406A here and I am just sending a signal into the ADC stage. So the phase noise performance will presumably be limited to about -136dBc/Hz by the 14bit ADC and the DSP after it.Note that the screen says Err in red (Error) because I've broken (bypassed) the RF signal path inside the analyser and so it has failed one of its automated signal path health checks in the RF converter section. The error goes away once I restore the full RF signal path to the ADC.

author: uncle_bob
author_rank: Unknown
timestamp: April 23, 2016, 01:41:30 pm
content: HiAnother way to come up with the noise bandwidth is to put in a known level of noise into the analyzer. There an large number of ways to do this. They fall into a couple of groups:1) Use something like a noise diode and possibly an amplifier with known gain.2) Modulate a signal with a noise sourceBoth have their limitations.The first one relies on the noise diode being correct (calibrated) and the amp (if used) having known gain when noise loaded. You still have to calibrate all that against a reference into the analyzer. You then will start asking questions about the flatness of your SA's response. I know this because I use this approach to check how far out some of my gear is. It's always a disappointing thing to check. A couple of db is not at all unusual and that messes this up for really wide measurements.The second one appears to have all the problems of the first plus some of it's own. If you do it with analog gear, that pretty much is the case. You need all the calibrated noise stuff with flat response *plus* your generator needs to have a really good modulator. The way that's normally gotten around is to generate the signal digitally. That gets you right back to the same issues we have been looking at on ADC's except on DAC's. In most cases you can do a pretty good job. You then do a sweep run over the full bandwidth and ... yes indeed ... there's a pesky db or three hiding in the signal chain.Could you go crazy an take out the flatness? Sure you could. It's normally easier to use this as a narrow(er) band approach and avoid the issue. Because of all the details, it's still not a great way to settle a beer bet ... (I had to get that in here somehow ..). The bet in this case would be:Does my analyzer "throw away" noise with it's FFT windowing process or does it all show up in a bucket somewhere somehow?Hint: This is why Matlab exists ...Bob

author: Performa01
author_rank: Super Contributor
timestamp: April 24, 2016, 09:39:53 am
content: Quote from: G0HZU on April 21, 2016, 03:38:16 pmWell, that’s a good suggestion and I’d certainly give it a try one day.Last time I’ve produced my own homemade PCBs has been more than 20 years ago and I guess the equipment is crusty by now and my stock of photoresist boards would be long dead. They are just FR4 anyway. So I’d have to order new photoresist boards and chemistry first at least.In short, right now it would be far too much hassle tinkering a PCB - and ordering a professional made PCB just for prototyping a circuit with a hand full of components appears odd (and expensive).The awful aspect of GHz circuits is the fact that we cannot just botch something together on a veroboard – all the more so if we want to use stripline resonators.But then, for evaluating an SA, I don’t need a super clean oscillator anyway and I’m pretty sure I can get away with what I have on hand. Might post some more results later…

author: Performa01
author_rank: Super Contributor
timestamp: April 24, 2016, 09:54:56 am
content: Quote from: G0HZU on April 22, 2016, 12:18:29 pmThat’s an impressive plot from an impressive instrument!Sure I’d be interested in your design and particularly the comparison of simulation vs the real deal would be very instructive for everyone.

author: Performa01
author_rank: Super Contributor
timestamp: April 24, 2016, 10:18:51 am
content: Quote from: G0HZU on April 23, 2016, 11:32:00 amWell, you’re certainly right that -121dBc at 2MHz is poor and there are indeed better ones available – and worse ones as well. The -125dBc at 500MHz or 1GHz is a different story though.The whole point is, as the SA fails to show a phase noise even remotely close to these figures, the signal generator is certainly good enough for these tests.QuoteYes, of course that’s always an option. It’s basically what we have to do every now and then (way too often) – use some external devices or even tinker something together in order to overcome some limitation in a test instrument.

author: Performa01
author_rank: Super Contributor
timestamp: April 24, 2016, 11:55:21 am
content: 2MHz Phase Noise SA44I’d like to show some closer inspection on the phase noise of the SA44.There are strong hints on local oscillator PLL phase noise limiting the dynamic range in closer proximity of the carrier. But the SA also uses the flat-top FFT window, which gives the best amplitude accuracy but isn’t particularly great in terms of side lobe suppression.So let’s start over again with the 2MHz signal at -15dBm, whose phase noise never exceeds -121dBc/Hz at distances >10kHz from the carrier, as we have verified by now. Span is 500kHz and RBW 100Hz and we get the familiar picture (SA44_RBW100Hz_2MHz_-15dBm_avg4)Phase noise is some -88dBc at 100Hz RBW and this would roughly mean -108dBc/Hz. Once again the abrupt step some 130kHz to the right from the carrier is striking. Of course this exists to the left as well, just not visible in that image above.Now this SA has a nice utility function to produce a phase noise plot: In order to use this, the span has to be lowered to a maximum of 10kHz first. So I set a 4kHz span and 6.5Hz RBW (SA44_RBW6Hz_2MHz_-15dBm_avg8)At that setting, we now clearly see all the sidebands from residual AM/FM, but also the phase noise which drops below -100dBc at about 1.3kHz distance from the carrier. So at least with very narrow resolution bandwidths, this SA is capable of providing >100dB dynamic range, but the phase noise from the signal source drops accordingly, so this doesn’t help with phase noise measurementsNow let’s start the plot (SA44_PhaseNoisePlot_2MHz_-15dBm_avg8)The phase noise plot shows nothing we could not already estimate from the first measurement at 500kHz span and 100Hz RBW – the phase noise at 200kHz distance from the carrier is displayed at about -109dBc/Hz, which is some 12dB worse than what the real value would be.Once again there’s a striking step at 200kHz distance from the carrier, where the displayed phase noise suddenly drops below -120dBc/Hz. It looks like we could indeed get some meaningful measurements at distances >200kHz from the carrier. But usually we are particularly interested in the range <100kHz, so this just means we might at least be able to measure the general wideband noise level of any signal source with reasonable accuracy.Below 3kHz, the phase noise increases a lot and it would be interesting to know whether it’s the signal generator or the SA44 dominating here. So looking at some FFT16 results again in order to determine the respective phase noise figures (FFT16_512k_2MHz_+3dBm_avg_zoom)Assuming 36.23Hz noise bandwidth from the 256k points FFT, we have to add 15.6dB to the measurements to get the phase noise normalized. The following table compares the normalized noise levels from the FFT16 measurements and the SA44 noise plot:FrequencyFFT16 Phase NoiseSA44 Phase Noise100Hz-61dBc/Hz-56dBc/Hz200Hz-71dBc/Hz-67dBc/Hz500Hz-88dBc/Hz-88dBc/Hz1kHz-103dBc/Hz-101dBc/Hz2kHz-114dBc/Hz-110dBc/Hz5kHz-120dBc/Hz-113dBc/HzIt appears the SA44 is not so bad here, giving results only slightly worse than the FFT16 measurements and being even in agreement at 500Hz.

author: Performa01
author_rank: Super Contributor
timestamp: April 24, 2016, 12:18:45 pm
content: 1-4GHz Phase Noise SA44I’ve already shown the spectrum of a 1GHz signal, just for completeness here it is once again, this time at a level of -40dBm with preamplifier on and 100Hz RBW (SA44_RBW100Hz_1GHz_-40dBm_P_avg8)Worst case phase noise is about -80dBc at 100Hz RBW or -100dBc/Hz.The phase noise plot looks slightly worse than the picture above, but close enough in my book (SA44_PhaseNoisePlot_1GHz_-30dBm_P_avg8)Now the same measurements at 3GHz (SA44_RBW100Hz_3GHz_-40dBm_P_avg8)Phase noise plot at 3GHz (SA44_PhaseNoisePlot_3GHz_-40dBm_P_avg8)Phase noise at 3GHz is about 10dB worse than at 1GHz. So let’s finally look at 4GHz (SA44_RBW100Hz_4GHz_-40dBm_P_avg8)…and the phase noise plot (SA44_PhaseNoisePlot_4GHz_-40dBm_P_avg8)Once again, the phase noise got a little worse, but only by some 2dB.When comparing this with the noise plot shown in reply #36 by G0HZU, it appears that my test setup performs equally well between 300Hz and 3kHz – and even better below 300Hz. Above 3kHz, the combination of free-running oscillator and Agilent signal analyzer pulls away…I guess all this finally confirms that it’s the LO noise we’re seeing, as FFT noise wouldn’t depend on the signal frequency. Still not too bad and very usable in my book – especially for the price.Particularly, if compared to the Agilent E7495B as seen in this video at 17:23:I tried to resemble the settings: 1.35GHz signal at -10dBm, 100MHz span, 100kHz RBW (SA44_RBW100Hz_1350MHz_-10dBm_P_avg8)The SNR is about 73dB, whereas it is only some 65dB on the E7495B. Of course, following uncle Bob’s best practice, I’ve cheated a bit. The reference level is set at 0dBm because there really is no need for 30dB headroom above the signal. This means less internal attenuation (which is unknown on the E7495B because it doesn’t show) and lower noise floor. Also the vertical scale was left at 10dB/div, which doesn’t change the SNR but makes it look better visually.

author: G0HZU
author_rank: Super Contributor
timestamp: April 24, 2016, 02:35:17 pm
content: See below for a screenshot of the basic simulation for the 4.4GHz oscillator. This is just an 'old school' small signal open loop analysis and there are better ways to do this.But the goal here was just to make something that worked. I know from experience that the oscillation frequency is always a few percent lower then the small signal prediction so it was design centred for 4.6GHz. In reality, it oscillates at 4.45GHz.The simulation is done using Agilent Genesys and the Sonnet EM engine. The PCB layout is simulated along with the components and the simulation helps optimise the layout to make sure the zero phase point aligns with the peak in loaded Q (or peak in group delay)See also a link to a youtube video showing the short term stability on my old HP8566B spectrum analyser. The phase noise on this analyser is fairly similar to the oscillator so it can't measure the phase noise as accurately as the E5052A analyser at work. Also I turned down the bias to 60mA to improve the short term stability. The oscillator needs to be kept away from changing air currents and vibration so I've put a plastic cover over it during the stability test.But you can see how stable it is. The CF is 4.455GHz and the span is only 100kHz and the oscillator only drifts a few kHz in a minute. The expensive PCB material helps here and I also used ATC 600S caps.The video is a bit dull as it just shows the analyser display for about a minute but then the camera pans and zooms to show the oscillator at the end of the video. You can see the oscillator jump around in frequency as I operate the pan and zoom on the camera. It only takes the slightest bit of vibration or changes in air current/temperature to cause the oscillator to jump around. It really needs to be in a metal/screened box...https://youtu.be/PMZyMLC2swI

author: uncle_bob
author_rank: Unknown
timestamp: April 24, 2016, 05:51:58 pm
content: Quote from: G0HZU on April 24, 2016, 02:35:17 pmHiA (dry) bar towel works wonders for slowing down the air induced wander in an oscillator breadboard ....If none is available, a spare ESD lab smock can be substituted.Bob

author: KE5FX
author_rank: Super Contributor
timestamp: June 14, 2016, 09:38:39 pm
content: Quote from: G0HZU on April 24, 2016, 02:35:17 pmThis is extremely cool. I wouldn't have guessed you could get -96 at 10 kHz out of a free-running oscillator at 4 GHz like this without going to a lot more trouble. For instance, I'm surprised that the surface roughness of the solder on your resonator isn't degrading it quite a bit.

author: G0HZU
author_rank: Super Contributor
timestamp: June 14, 2016, 11:35:16 pm
content: Quote from: KE5FX on June 14, 2016, 09:38:39 pmYes, you can see that the PCB has seen a bit of action with the soldering iron. When it was at work in front of the E5052A I couldn't resist experimenting with different caps and also I rotated the caps slightly to try with different tap points into the resonator. So what you see is the aftermath of this. You can also see in places how flat the solder 'was' before I did all this because it was originally flattened with solder wick.But you can still see that the noise on the HP8566B is still in the right ballpark. The HP8566B has similar/worse phase noise than the oscillator up here.QuoteYou can see in my image that I've used desoldering braid to keep it looking tidy but this was before I messed with the caps at work.What I can say from experience is that any attempt to use sticky copper tape instead of the original PCB copper for the resonator does not work well at all.I've got a similar oscillator design somewhere here that runs at 6.7GHz and it had similar phase noise performance to the one at 4.4GHz. I've built quite a few oscillators like this and these were used (for example) to test the noise performance of the first few ranges on my HP8566B analyser. The main aim was to make something quick, simple and stable with similar phase noise to the HP8566B.

author: KE5FX
author_rank: Super Contributor
timestamp: June 15, 2016, 02:22:15 am
content: Quote from: G0HZU on June 14, 2016, 11:35:16 pmI don't know if you subscribe to QEX or not, but this might be a good thing to write up for them.

author: radar_macgyver
author_rank: Frequent Contributor
timestamp: June 15, 2016, 06:27:04 am
content: Quote from: Performa01 on April 24, 2016, 12:18:45 pmThose wiggles < 300 Hz could be since you're running off the internal reference, which in an SA44 is a TCXO. Locking to an external stable reference would help get rid of them.This postby hendorog links to an article by Signal Hound where they try to characterize the phase noise of an SA44. You can clearly see the effects of an improved reference oscillator.Thanks for the writeup. I'll send a link to this thread next time someone brings this up at workI have a hard time explaining why we still need 16-bit A/Ds for receiver front-ends when folks will pull out a data sheet for the latest whizbang 10 GSPS 8-bit A/D and say "we can just filter it with an FPGA".

author: hendorog
author_rank: Super Contributor
timestamp: June 16, 2016, 07:33:54 am
content: Quote from: Performa01 on April 24, 2016, 10:18:51 amAccording to SH:"It’s actually a sigma delta 24 bit ADC, digitally converted to 16-bit I&Q."https://signalhound.com/support/forums/topic/sa44b-adc-bits-resolution/

author: uncle_bob
author_rank: Unknown
timestamp: June 16, 2016, 07:47:22 pm
content: Quote from: hendorog on June 16, 2016, 07:33:54 amHi..... which seems like a really silly thing to do if you actually *have* a 24 bit ADC ...Bob

author: hendorog
author_rank: Super Contributor
timestamp: June 16, 2016, 08:06:07 pm
content: Quote from: uncle_bob on June 16, 2016, 07:47:22 pmIf that guy said they do, then they do. He is one of the engineers not some marketing rooster.The ADC was questioned to be 14bits, turns out its 24bits operating effectively at 16bits. Thats a good thing, not a bad thingIts really only silly if they then spend effort digitally converting back to something greater than 16 bits again isn't it?

author: uncle_bob
author_rank: Unknown
timestamp: June 16, 2016, 08:11:42 pm
content: Quote from: hendorog on June 16, 2016, 08:06:07 pmHiI can go out and buy a part that is indeed a 24 bit ADC. The data sheet says 24 bit ADC and at *some* clock rate it operates as a 24 bit ADC. If somebody asks me a question about "what sort of ADC?", I can honestly answer 24 bit ADC.The gotcha is: It's a part that puts out 24 bits at a 15 Hz sample rate. At a higher rate, it puts out fewer bits. If I'm using it at 100 KHz, it might only be a 12 bit ADC. Since it's only 12 bits, running the rest of the system at 16 is fine.Bob

author: hendorog
author_rank: Super Contributor
timestamp: June 16, 2016, 08:23:27 pm
content: Quote from: uncle_bob on June 16, 2016, 08:11:42 pmYep I understand that - the way you worded your first comment it sounded like you were saying they were lying about the chip being there in the first place.There is an implication that its not that simple though - I mean just running the ADC at 16bits doesn't really qualify as 'digitally converted' does it?It would be interesting to know more about how they do it. (Edit: I asked the question in the SH forum so will see what they say)

author: uncle_bob
author_rank: Unknown
timestamp: June 16, 2016, 08:40:05 pm
content: Quote from: hendorog on June 16, 2016, 08:23:27 pmHiIf I spend the extra money to buy an ADC that really does a legit 24 bits at a 240 to 480 KHz sample rate, I've already put some money into the design. Bumping the FPGA up one notch to handle 24 bits rather than 16 bits is *way* less money than what I just paid to go from 16 to 24 bits on the ADC. It does not seem like a logical decision for a designer to make. It sounds a lot more like a marketing guy coming up with neat headlines. There is no reason to toss out 8 bits just to get to I/Q samples....Some basic math:16 bit ADC <$1024 bit ADC >$4016 bit capable FPGA ~$12Same FPGA good to 24 bits ~$18(yes, there are some assumptions in that math)Bob

author: hendorog
author_rank: Super Contributor
timestamp: June 16, 2016, 09:53:01 pm
content: Quote from: uncle_bob on June 16, 2016, 08:40:05 pmWell maybe you are right, you have a lot of knowledge and experience. However I can't help but think you are jumping to conclusions:Firstly, it isn't in any marketing I have seen. The device has been out for years and this recent post about bit depth is in response to a question which was asked. The marketing says '16 bits'Secondly, this is an inexpensive device. Its been well documented that they have built in some LO switching cleverness to remove spurious and keep the hardware price down. So why would they push the price up by adding a part that isn't required?

author: uncle_bob
author_rank: Unknown
timestamp: June 16, 2016, 09:55:41 pm
content: Quote from: hendorog on June 16, 2016, 09:53:01 pmHiExactly my point: Why would they spend a lot of money on an expensive 24 bit ADC (they are expensive) and then not use it? They could save a lot of money just buying the lower cost 16 bit instead.Bob

author: KE5FX
author_rank: Super Contributor
timestamp: June 16, 2016, 10:07:09 pm
content: The guy in question knows what he's doing, so he must've had a reason that isn't apparent from the outside looking in. For instance, maybe they just stumbled on an unusually good deal on a bunch of those 24-bit converters on eBay.In low-to-medium volume production with aggressive cost targets, part selection is as much an input to engineering as it is an output.There could be purely technical reasons as well. Antialiasing requirements are somewhat relaxed with sigma-delta converters since they oversample by nature, and that's a win because filters are always a hassle. Especially if you're designing a spectrum analyzer, where you'd like to achieve spur rejection in the vicinity of 100 dBc or better. So either he spends a lot of money buying ready-made LPFs from Mini-Circuits or whoever, or he spends a lot of board space (likely with accompanying shielding requirements) on discrete inductors and capacitors. Unintuitive choices can make more sense when you're simultaneously fighting for space, power, and cost like they are.

author: uncle_bob
author_rank: Unknown
timestamp: June 16, 2016, 10:09:32 pm
content: HiTo be very clear, I'm not saying anybody is lying. I'm just saying that, on the face of it, it does not make any sense.Bob

author: hendorog
author_rank: Super Contributor
timestamp: June 17, 2016, 01:38:33 am
content: Quote from: uncle_bob on June 16, 2016, 10:09:32 pmYep understood. Hopefully Justin replies to my question and we can find out more about it. Like KE5FX says there must be a good reason for it.Maybe I'll even get some beer back

author: bson
author_rank: Unknown
timestamp: June 17, 2016, 01:51:48 am
content: The AD9864 appears to be a highly integrated digital IF subsystem and the ADC itself is just a part of it. Perhaps it's simply not available in a 16-bit version?

author: uncle_bob
author_rank: Unknown
timestamp: June 17, 2016, 02:23:31 am
content: Quote from: bson on June 17, 2016, 01:51:48 amHiThe sigma delta in the 9864 is exactly the sort of part that I was talking about. At reasonable data rates it is a 16 bit converter. As you decimate the output, you gain bits through the filtering process. At very low clock rates you fill up more of the i/o word with information (as opposed to noise). At the full rates they advertise for the Signal Hound, it's a 16 bit part. In very narrow bandwidths it is a bit better than that.Simply put, sure, it's a 24 bit part in some cases. No it's not a 24 bit part the way they use it. The way they use it, it's a 16 bit part. Thus the 16 bits into the processing end of things. Makes perfect sense.Bob