Designing matching network to drive switching FET for HF amplifier - equivalent? - Page 1
Forum section: electronics / rf-microwave

author: wb0gaz
author_rank: Frequent Contributor
timestamp: March 26, 2021, 09:25:36 pm
content: I'm interested in designing a (band-specific) matching network that would transform 50 ohms to the input impedance of a switching FET (for example, IRF510).The IRF510 (as seems to be the case for many switching FETs) specifies input capacitance as a fixed value (i.e., 170 pF).Can I use this value (plus a small series resistance I'd use at the gate of the FET) as the impedance to be matched (to 50 ohms, for the band of interest)?For experimental verification, can I then use a capacitor (instead of the FET, with the small series resistance added) to verify the behavior of the matching network?My area of interest is HF (1.5-30 MHz).Thanks for any suggestions, pointers, or requests for clarifying my question!Dave

author: T3sl4co1l
author_rank: Super Contributor
timestamp: March 26, 2021, 09:37:50 pm
content: Not quite, it's a bit more than that, also you have to include Miller effect if it's common source. The resistance is substantial (some ohms?), and I think has a dominant diffusion component in that range (so, expect gain to fall off as ~1/sqrt(f) up there).You can get pretty close already with just 50 ohms into a termination (Thevenin eq 25 ohms, good to 32MHz into 200pF, and say 30MHz into 300 maybe 400pF with a 2-3 order peaking network), and for a resistive input impedance just ballast it (with Zobels or whatever) as needed.To go faster (or to deal with substantial Miller effect), just scale everything down by impedance, put a transformer in front, and accept the lower gain.It's not a 1:1 gain-bandwidth tradeoff, because of that diffusion component, at least on most old MOSFETs where that applies; effectively, GBW rises as gain falls. But the returns diminish rapidly, when the alternative is, you could just plop in a 2N3866 and get say 10 times the GBW, and the same power output, for 1/10th the supply current.I didn't characterize IRF510 very much, but I have looked at 2N7002 cascode, and found these sorts of conditions. That one happens to be good for over 200MHz, but only if you don't mind that the gain is like 2 or so; and with the, whatever I did I forget, 200-400mA or something like that, supply current? Yeah, not a great deal.Tim

author: wb0gaz
author_rank: Frequent Contributor
timestamp: March 27, 2021, 12:55:22 am
content: Thanks very much, Tim --- like my previous project (LC filter) this is as much a learning experiment as anything else (and fortunately had a very good result after starting out "in the weeds")To clarify my intention with this project:1. I plan 4:1 or 9:1 broadband input transformer.2. My target devices are eventually IRFP360LC; for those the gate capacitance would be 2200 pF or thereabouts each; it seems that just driving a 50 (or whatever) ohm load resistor with the gate across it (AC coupled, net of bias path) would be "difficult". I'm primarily interested in lower HF bands (1.8-7.3 MHz.) I expect to have a separate input matching network for each band, as well as attenuation to equalize gain across bands.3. My mention of IRF510 was as a getting-started experiment (to see what happens at lower power levels), and I have a bunch of those (less voltage/current = less collateral damage *when* I blow them up).4. I plan initially to try AB1 class operation (vs. Class E), just to get a feel for "do it yourself" results - bias up to a point of idling current, then apply drive.5. If i were to disregard miller effect initially, would the match be a starting point for a very low drive level (say, 30 dB below normal drive level)?6. I plan use a current-limited DC drain supply.Thanks again for your initial comments - as before, you're helping me get started!Dave

author: T3sl4co1l
author_rank: Super Contributor
timestamp: March 27, 2021, 09:36:20 am
content: I made this a while ago,Still needs better biasing (temp compensation), but it does what it says on the tin, overall what was it 40dB gain?, 4W, though not clean up at 50MHz, the slew rate from those 47 ohm pulldowns (R10, R23) isn't enough. Because of course, the gates move a bit slower than needed for full power. Lacking complements for Q2/Q6, or higher current types, it's the best I could do from salvaged parts.Also, the transformers are all matched transmission lines, overkill for this bandwidth but a nice exercise. They're probably flat to well past what the amplifier is capable of. (Note L1 is a dummy delay line, not so much an actual transformer. Again, hardly necessary.)Note, traditional amplifier design doesn't simply stop working at some frequency; while you wouldn't be able to put global negative feedback around this because of phase shift, local feedback is fine (indeed, see the source degeneration), and it's not like Q2/Q6 have matched impedance environments on either side, the emitter follower is doing what it's supposed to do* (if just not with enough pull-down current for full signal swing). The impedance there, is just whatever the emitter and gate happen to be.*Which isn't always the case at high enough frequencies; fortunately these are rather fast transistors (2GHz!) so they follow well. Up near fT, emitter followers stop followering so well, and the impedances go wonky (emitter goes inductive, base goes ~90 degrees from emitter load impedance, which is to say there's more feedback from output to input so it's a poor isolator/buffer, etc.).Regarding (5), match doesn't matter much with signal level; well, it does vary at high level, but I think most times not enough to matter more than, like, one rev to adjust it and be done. But yeah, small signal testing is fine, bias it up and see. You can always test return loss, characterize it that way. That's what I did, as well as s21 etc.Oh, I also did a higher voltage power transistor, with a 3:1 gate transformer...https://www.eevblog.com/forum/rf-microwave/superjunction-mosfets-for-rf/The transformer is three transmission lines of "star quad". Two are on the pot core, one is in free space (with a blue ferrite bead just for kicks). The primary ends are all wired in series, so the characteristic impedance at the input is more like 75 or 80 ohms; close enough. The secondary ends are all wired in parallel, to a flat strip of very thin PCB stock, which goes over to the gate terminal. So, very very low drive impedance, about 5.6 ohms. This pushes around the gate pretty well I think, and also matches well over a modest frequency range.Star quad is 4 wires twisted, with opposing wires connected in parallel. This gives about half the characteristic impedance of two separate twisted pairs in parallel (~50 ohm), so it's a very efficient way to get low impedances with wires.The "start" ends of two of the transmission lines have common mode voltage, hence the pot core; of the three in series, the bottom has none (so doesn't need to go through the core), the middle one sees 1/3 the signal voltage and the top one sees 2/3. So the middle one makes 2 turns, and the top one makes 4 turns, on the pot core.The rest is, let's see, I think:Active device: STF6N60M2Gate coupling: from autoformer, 2 x 1uF 0805 in parallelDC bias: 10kR to trimpotTrimmer regulated with a 12V zener, and 47k pullup to B+.Source coupling: 2 x 1uF (leaded MLCC) || 0.1uF (PP film) || 2 x 51R 0.5W to GNDSo, ample DC degeneration, which explains the relatively high gate bias. Won't be terribly unstable. (I recall it did tend to drift a lot in testing, and a fan helped. I mean, that heatsink is in no position to do much good by itself, so that would happen.)Drain coupling: couple mH inductor (powdered iron core, 1A or so rated) up to B+, 47nF (film) to output (so, direct to 50 ohms).I'm not sure what the feedback measurement was, it doesn't make any sense for it to decrease with frequency? The others are fine though. Eh, the s11 is probably a bit dubious, that janky reflectance bridge isn't great above 30MHz, really.Anyway, the gain is consistent with a C-loaded Norton equivalent source (drain current || Coss || R_L), and with gain dropping more or less as sqrt(f). I'd need to repeat the s11 measurement with a better reflectance bridge to have a better measure of gate impedance.The funny thing is, these transistors are more than happy to oscillate at 300, 400MHz, when the gate is capacitively loaded -- the amplitude isn't much, though -- I mean, more than enough to handily fail emissions in a SMPS, but still a fraction of the full operating range. Don't let that number fool you, though: the important part is the bandwidth of that oscillation (its envelope rise/fall time), which is as expected, mid 10s of MHz. It's the bandwidth that you can actually use, whether starting from zero as here (50 ohm load, baseband / untuned), or tuned (hmm, might not be very powerful at 100MHz, but should be pretty effective in the SW band, say?). And it should have enough bandwidth to, say, cover the entire FM BCB if you wanted to power-amp the whole thing for some reason, or most or all of the SW band at once.Tim