VFD Output filter - Page 1
Forum section: electronics / projects

author: Floyo
author_rank: Regular Contributor
timestamp: July 16, 2017, 09:49:05 am
content: Hi All,I recently bought a lathe with a 3 phase motor, and I only have a 230v single phase hookup,so VFD it is. However since the lathe is from the mid 70s and has a somewhat custom motor I want to makesure the winding insulation and bearings don't get hammered too much by the fast edges from the VFD.Since the motor and my lab do not occupy the same physical space I have been using a bunch of incandescent bulbs as loads,not the same by far, but it will have to do for now. The power level is pretty low in the grand scheme of things with a0.55Kw drive, maximum current is 2.6A as per the spec.The "DV/DT" differential mode filter wasn't too much of a challenge, some normal ferriteoff the shelf inductors from the parts bin work fine, Powdered metal cores (Micrometals T106-26) gave too much core losses for their size.Some damping resistance stops the filter from oscillating and gives pretty decent edges with a more controlled rise-time.However, the common mode filter is where the issues begin, since the common mode currents are quite high (200mA-1A),and all cores I had available saturate at just a couple of mA. Thus providing no useful attenuation/impedance at that point.I did try a normal ferrite core, which fairs much better, but due to the lower permeability the inductance of courselacks a fair bit. Note that the mains earth terminal of the drive is connected via a 220n cap to the positive DC bus voltage internally, providing the return path for the common mode current.Reading some papers and some preliminary research pointed me to some common mode cores from Vacuumschmelze that aremade to handle this situation better, but those too will saturate quite soon with a handful of turns on them.http://www.vacuumschmelze.com/en/products/cores-components/applications/cores/vitroperm-cores-vp-250-f.htmlSo long story short, how would you handle this situation, use the "proper" cores, or just a plain old switching ferrite with an air gap, etc.Keep in mind this is a hobby project, so actually passing any safety/EMC requirements is not part of the problem, but getting a "good enough"solution is. I am not a complete stranger to SMPS/magnetics design, but this 3 phase motor drive stuff is new to me, so any good background info would also be much appreciated, as I am not quite sure on where to start on a proper design yet.Attached is the rough schematic of the test setup.

author: jbb
author_rank: Super Contributor
timestamp: July 16, 2017, 10:08:51 am
content: Good on you for thinking of the motor.For safety, the most important thing is to maintan a nice solid earth connection.For the CM choke, you could try adding a 4th winding with a load resistor, I.e turn it into a transformer. Not sure how effective it will be, but worth a shot. Also be careful to bring the earth wire from your filter straight back to the VFD so you don't make a big ground loop.As it comes to configuring the VFD, the advanced vector control methods may be confused by the LCL filter, so if it acts weird, you could try the old fashioned constant flux (aka constant Volts per Herz) mode.

author: SeanB
author_rank: Super Contributor
timestamp: July 16, 2017, 10:40:51 am
content: Just run it as is, without the filter. 0.55kW is a small motor, and use a shielded mains cable from the VFD to the motor, and connect shield to motor frame and VFD ground connection, as in the manual. I am running a Lenze motor from 1970 with that, and it has been fine running with that for close to a decade, with no issues other than me adding an external fan instead of the existing enclosed fan, as I wanted it to slow down to 40Hz to replace the dead varispeed belt drive it had that had shattered it's spring steel disk. I would however add in an incoming mains filter, and set the inverter to drive it with a linear V/Hz ratio, not the square law it sometimes uses for fan loads.

author: T3sl4co1l
author_rank: Super Contributor
timestamp: July 16, 2017, 02:08:12 pm
content: Quote from: Floyo on July 16, 2017, 09:49:05 amWell, fine, the CM current into a low impedance is 200mA+, but what's the voltage into a high impedance? If it's unlimited voltage (ideal CCS), then sure: you have no hope, it will saturate anything.I'm guessing the voltage is very much finite, and you need only find a CMC that handles enough flux. Then the current will stay low.QuoteUhh...This thing isn't on a GFCI / RCD circuit, is it...220n isWAYtoo much ground leakage for a mains operated device. 4.7n is typically all that's used in that location.220n to just the filter is okay -- if you need that much (try getting away with less?). Then you'll probably need an additional filter outside of that one, to clean up with respect to PE (and therefore motor frame).Another idea that may help: try filtering each line independently. All three have a low (RF) impedance to DC bus, and the RF is probably not balanced three phase, so there's no point in filtering from phase to phase, as such. This puts most of the RF on the filter chokes, so that there won't be nearly as much to deal with in common mode.QuoteYes, they are very good. The flux handling (per turn) is higher (because the saturation flux density is higher), and the residual current (that is, the magnetizing current due to I = Vcm / Zchoke) is smaller (A_L is larger). They are rather pricey, though!QuoteOh. Well, "good enough" is probably nothing at all. Stuff it in metallic conduit so it doesn't blot out radio stations, and don't put the VFD terribly far from the motor. Fine with me...I suppose if you need to get every last watt from that motor, an RF filter will reduce core losses. It's still a lot of trouble to go to, though, for not much gain (~10%?).Tim

author: Floyo
author_rank: Regular Contributor
timestamp: July 16, 2017, 07:24:07 pm
content: The open circuit voltage is indeed not infinite, but around 230V RMS, with a low frequencycomponent with the HF crud superimposed, screenshots attached.The yellow trace is the voltage between the DM filter capacitors starpoint, and mains earth.The current trace (blue) is the common mode current measured by taking together all three phase wires in the current probe,in this test no CM filtering was present.To clarify, the 220n cap is not a classical y cap from line to earth, its from the DCbus to earth, and wasput there by the manufacturer, not me :p.The input common mode filter of course also has a y cap, which at 22n is substantial, but still fineon a 30mA GFCI.I'm not sure I quite get how you envision the independent filter per line though.As for the cost and the "good enough", I'm aware just hooking up the motor with some short shielded cablewill most likely be just fine, but I'd rather be sure since the motors are pretty rare and expensive, soa few tens of euro's on parts and some time is no big deal. Furthermore its a good learning experience towork out all the magnetics etc.But that would then leave the design of the CM transformer, and I'm still a little lost in the magneticsjungle on how to proceed. So I have a core with a certain saturation density (T) and magnetic cross section (m^2).Then there is the V*s product as the design input which will dictate whether the core will saturate or not.But I guess first would be setting some kind of goal for the needed attenuation. Lets set 20dB/10x as a nice round (low) number.a 20n cap would have a 500Ohm impedance at the 16kHz fundamental frequency. So that means a 5Kohm common mode impedance is neededat 16Khz, so 50mH of inductance. The current would then be 46mA.Then taking the 50mH*46mA = 23mWb of magnetic flux.Now lets take the T60006-L2030-W981 VAC core, with 0,40cm^2 cross section and 1,2T saturation flux density. This thus gives48µWb of available flux.But this is around where I start getting lost in the woods :p, since somewhere in this the number of turns, the magnetic path length and the fixed Al value of a toroidall still need to come into play, not to mention the fact that all this assumes the signals are sinusoidal, which they are obviously not.So any help in figuring this out would be greatly appreciated.

author: T3sl4co1l
author_rank: Super Contributor
timestamp: July 17, 2017, 02:17:04 am
content: Quote from: Floyo on July 16, 2017, 07:24:07 pmHmm, quite low frequency (16kHz). A lot of flux indeed...Note that that applies to the DM chokes as well, but they also have to handle peak load current too, so whatever, they'll figure it out.You do have the option of rounding off the edges (which is your concern as far as whipping voltages around), not filtering the whole thing. Your initial choice of f0 ~ 230kHz seems to support this.Yeah, just put the CMC after the DM filter. Don't try to filter what you don't need to.QuoteWow. That feels... really irresponsible.A single point of failure, between DC bus and exploding fuses?QuoteYeah, but, DC bus has full mains ripple on it too, so...Unless this is 3-phase input, in which case, the ripple is, I forget exactly, around 10% I think. Still too much for a 220n cap though.QuoteInstead of thinking of it as filtering from line-to-line, filter line-to-ground. That's all.QuoteHold on here:Flux and current are two different things: they come from B and H respectively. B and H are related by mu, which means a high permeability material has small H inside it (and consequently, will give small I and large Z in circuit).If you take a ferrite EE core and hold it around a bobbin, while pulsing the winding (on the bobbin) with a transistor, you can measure the saturation point as you move the core around. (That is, the transistor turns on, dumping a constant voltage across the winding. A shunt measures inductor current, and at some point in time, current starts rising steeply: saturation.) What you find is: as you adjust the core gap, the initial slope changes, proportionally; but the saturation time does not change*.*It does actually change, but only a little, not proportionally. The reason is due to voltage drop in the winding, thus taking more time to reach saturation. The core also needs to be deeper into saturation before saturation becomes visible (i.e., before effective mu drops).(Effective mu is the permeability you get, assuming the gap is distributed. Or, to think of it another way: the magnetic path length includes l_e through the core, plus l_g through the gap. But the equivalent air-gap length of the core is l_e / mu_r. So the equivalent total airgap is l_g + l_e / mu_r. Going back to permeability, this gives:mu_eff = mu_r * l_e / (l_g * mu + l_e))The reason is obvious, when you think about it: the core cross section remains the same, as does flux density at saturation. Applied voltage and on-time remain constant (i.e., applied flux), so the saturation time also remains constant. What you're changing is the amount of current drawn by that flux.For gapped ferrite cores, it's best to start with flux and area, then design the gap based on current and inductance.For fixed-geometry cores (pregapped ferrites, toroids, powdered iron (distributed gap) cores), you don't have a choice, and either starting point is equivalent.But, keep in mind, when the permeability is really high, the tolerance is really wide, too (ferrites are ~15% or worse!). So for high-mu cores, it's still better to work with flux first. Then you get magnetizing current as an output, and you can decide if you need more turns to achieve less.QuoteThese calculations are all right, by the way. You're on the right track.(Actually, you're on the left track?But going the right way!)Going backwards to your starting point: 23mWb at 16kHz, square wave, is... 1472V.This comes from:Phi = Vpk / (4 * F)for a square wave. Change 4 --> 4.44 for Vrms sine.Well, that's definitely a conservative figure.Also, that multilevel waveform will probably take somewhat less flux. But this is okay for a conservative figure, too.If we crank that down to the minimum, i.e., about 150V, that gives an expected magnetizing current of 0.45A.That doesn't smell right. Those cores saturate with 10s of mA......Ah, Vitroperm 250F is the lower permeability material, not the top-notch stuff. Then, you won't have much advantage over ferrite, and yeah, your limit will be set more by magnetizing current than by flux.Having noted this, yeah, your numbers will work, and give the impedance you wanted -- unfortunately, you need (23mWb) / (48uWb) = 480 turns to get there!Also, A_L is stated as 2.4uH / t^2, which only needs 144 turns to get 50mH. Something else is fishy.If you went with peak, rather than peak-to-peak flux, that would account for a factor of 2, but still not for that 1-2/3 factor. Unsure.QuoteYou were very close: you had winding flux (Wb) and core flux (Wb/t), and only needed to divide them to get turns. A_L is the only figure with a square on a unit, everything else is straight proportional (well, okay, area is in m^2 and whatever multiplier you like, but you use areas for areas so that's fine..).Tim

author: tecman
author_rank: Frequent Contributor
timestamp: July 18, 2017, 05:05:38 pm
content: You should provide a good ground path for the rotor, to prevent induced current from damaging the bearings. This is mostly due to the high frequency PWM resulting in some common mode current from leakage flux. There are a number of commercial products, but most use a metallic brush to ground the shaft. Generally the voltage you will see on a shaft is low but the current spikes can be high (and damage the bearing).paul

author: Floyo
author_rank: Regular Contributor
timestamp: July 18, 2017, 07:38:56 pm
content: @tecman, Yeah, that phenomenon is the whole reason for me investigating this. If the filtering turns outnot to work out I can always try and see if a mechanical solution can be figured out, I do own a lathe after all.@TimThanks for the detailed responses.I rechecked the numbers, and I think the discrepancy in the number of turns came from the fact I was using both half the flux swing,and I used the full 1.2T in the calculations, whereas VAC specs their Al at 70% of that. Accounting for this makes the numbers come out prettyclose.Now, the number of turns will have a snowballs chance in hell of fitting on that core, especially in triplet, but those numberswere more about getting the units/math in order. The resonant frequency of the filter should be a fair bitabove the fundamental switching frequency, or the filter will want to oscillate a bit too readily.That significantly lowers the needed inductance, and makes the whole design a bit more attainable.A proper sine filter is out of the question, stuff just becomes too unwieldy.Two questions remain however, firstly, why is there an apparent discrepancy between the V*s and H*A approach?It seems the H*A figure is way overkill, but somehow that can't really be the case, right? Though I guessif you go by the V*s approach you'll just end up with less inductance in the end.Secondly, where does the 0.45A magnetizing current figure come from? It looks like the outcome of V=L*di/dtwith approx 300+V, 50mH and 16Khz, but wouldn't the cap prevent that current from ever building to such levelsin CM? And if its the DM current in the CM transformer, those fluxes should mostly cancel out and the core wont "notice" as such.I'm probably missing some obvious stuff, but its becoming clearer

author: T3sl4co1l
author_rank: Super Contributor
timestamp: July 19, 2017, 06:43:05 am
content: Quote from: Floyo on July 18, 2017, 07:38:56 pmOk, so the component values will likely be similar to your OP example.The filter will always oscillate, the question is: how much voltage and current do you want it to oscillate with?The amount of energy the filter has to deal with, is determined by its impedance. For a simple RLC series circuit, Q = Zo / R, and Zo = sqrt(L/C).Or for a typical ladder circuit, R comes from the source and/or load termination.If you pick a higher Zo, you get less peak current, but more voltage drop at lower frequencies.The motor will be a complex load at these frequencies, so the filter should be designed as a constant resistance or damped filter (or a diplexer with the high-pass port terminated).With (R||L)s, you've already got a head start on this. (R+C)s can also be used, to the same effect, and you might indeed consider using both. The resistors put zeroes in the response, so the attenuation flattens out at HF, rather than dropping asymptotically. You can follow this up with a proper lowpass, and then maybe more damping again, to get good attenuation and freedom from impedance changes.It can help to do, instead of C, or R+C, use C || (R+C). The single C is smaller, and provides asymptotic attenuation, while the R+C provides damping in the transition band.Analogously with the L (use L + (L || R)), but that's kind of annoying because more inductors.QuoteNot sure. In a linear material, that has accurately known parameters, it doesn't matter. Maybe that's the difference.QuoteIf the cap prevents it, why use so much in the first place? You'd of course want a complementary capacitance, so that the capacitor and inductor do all the filtering they can.If we make C = 4.7n, that's 2.1kohm at 16kHz. Fo will be higher than 16kHz then, and less voltage will drop across the inductor (most of the fundamental will pass through).That's not a good PWM filter. Minding that, to achieve a "good PWM filter", you'd need Fo more like 1kHz, which will need beefy inductors indeed... (Hey, I know where we can find inductors that big: inside the motor!...)So if we're not doing PWM filtering, what are we doing? Any other cutoff is arbitrary. 100kHz? 1MHz? 100MHz?At this point, you can just kind of throw some L and C at it (keeping Zo = sqrt(L/C) a reasonable value), and say "oh, yeah, it's filtering just fine", for the same arbitrariness of "fine".I mean -- actually; going back to my earlier statement that the motor windings are probably complex at high frequencies: youcanfind a basis for this cutoff, which aims to accomplish your original task (reduced peak voltages). But, you'll have to measure the motor winding impedances at high frequencies.Otherwise, you're just stabbing in the dark, and you can make rough hand-waving guesses like, that there are probably no strong impedance dips below 50 or 100kHz, and you should put your filter there, and dampen the transition band so as to be prepared for any impedance dips and peaks that might lie in wait.So; if you skip the first, most obvious condition (filtering PWM whole), the fallback is still reasonable (filter edges), but you need more information (motor Z(F)) to design it.Tim

author: T3sl4co1l
author_rank: Super Contributor
timestamp: July 19, 2017, 07:03:47 am
content: Some more comments on PWM filtering (okay to disregard):Normally, you set a PWM filter's Fo somewhere between the top of baseband bandwidth, and below a modest fraction of Fsw. Fo < Fsw gives some room for attenuation by Fsw.This puts the transition band (Fo give or take some factor) in a natural spectral gap, where no energy is present. Which keeps reactive power, and losses, small. The filter design is not very critical.For a class D audio amplifier, baseband is at least 20kHz, so Fo needs to be higher than that, and Fsw higher still. Fsw ~ 200kHz is very typical, and this is what puts it there. There is very little energy between 20kHz and 200kHz, basically just harmonics of the audio signal (which should be low, because it should be a low distortion amplifier), and maybe some weird quirks of Fsw itself (a clean oscillator won't have subharmonics, but a sigma-delta modulator will).If it's acceptable to have little attenuation at Fsw, you can place Fo above Fsw. But now, Fo is in the midst of a forest of harmonics and sidebands, and you have to worry about everything. There will be significant energy in the filter's transition band, and therefore it must be designed carefully. It needs a properly matched source and/or load, or damping, otherwise that energy piles up in the reactive components and you get bad behavior (huge input peak currents and output peak voltages, huge losses in L and C).You want to avoid putting a filter between, say, Fsw / 3 and Fsw * 5, because the fundamental and harmonics are strongest there. Higher starts to be okay again, because the harmonics go as 1/N (and power as 1/N^2), and while you still need proper termination, the amount of energy being absorbed in those terminations is more modest (single watts, not tens or hundreds).Tim

author: Floyo
author_rank: Regular Contributor
timestamp: July 19, 2017, 09:01:32 am
content: Yeah, hence where the rule of thumb of being 10X away from whatever it is you are trying to filter. Its a shame SiC (or GaN) vfds aren't really a thing yet, at 100Khz+ fsw the filters would become a lot more reasonable. On another note, (single phase) Vfds could really use a PFC on the input as well. I guess the industry in this regard moves pretty slowly, with modern drives still having the 16Khz fsw as the 20ish year old ones I have.Unfortunately I left my "network analyzer" (analog discovery) on the other side of the country, so I can't easily do any impedance measurements without taking half the lab with me. So I will just do some time domain measurements on the actual motor, and hook up the LCR meter for hopefully some kind of broad figures.

author: T3sl4co1l
author_rank: Super Contributor
timestamp: July 19, 2017, 09:58:01 am
content: A friend worked at Allen Bradley in the drives division; he said they are indeed very interested in pushing Fsw (or efficiency, because shorter commutation time per period) and density. But you aren't going to get that for nothing, you're going to pay for it. New tech, special specs, that kind of thing. If your experience is confined to bargain basement Chinese blocks, you're probably going to find an integrated IGBT bridge module, notoriously slow (hence the low kHz) but compact and cheap.Power electronics does tend to move slowly, particularly as you go up in power. 20+ year old designs are still in production, in the low megawatt ranges; in the power transmission sector (HVDC systems, say), they're largely SCR based, AFAIK. They still call the inverter buildings "valve houses", from their heritage using mercury thyratrons.Hybrid and electric automotive is probably the faster moving sector at those power levels (fractional megawatt), and a big driver in compactness.Tim

author: mikeg
author_rank: Newbie
timestamp: July 27, 2017, 07:09:26 pm
content: The issue is not the frequency, it's the edges. The super fast edges cause the problems with I leakage and insulation breakdown.What you want is a line reactor. A choke made for this. Search the web, you'll find many hits....For an old motor like this it may seem counter intuitive but you want a slow PWM freq with slow edges. The DI/DT issue is reduced.16Khz was chosen many years ago. Why: with MOSFETs there is a cross over frequency point between switching losses and conduction losses. Above this frequency switching losses dominate. Below this frequency conduction losses dominate. 16kHz is the magic number.I don't know if this is still true with the newer transistors but it has stuck with us.

author: floobydust
author_rank: Super Contributor
timestamp: July 27, 2017, 07:47:14 pm
content: I use off-the-shelf 3-phase output chokes (single EI core) because they have the voltage rating and safety approvals. I would not make one myself.PWM frequency is also limited by the motor's magnetics, grain-oriented steel.I find the motor winding stress (overshoot) caused by leakage (lead-in wiring) inductance.I've worked with 100HP motors on 10-50m cable runs. There can be some transmission-line tuning there, certain lengths/cable combinations make massive spikes so rewiring things stops IGBT and motor insulation failures.Then I started adding VFD output inductors to stop that hassle, and it costs more but works nicely for reliability.