How to design a transformer leakage for resonant converter? - Page 1
Forum section: electronics / projects

author: Miyuki
author_rank: Frequent Contributor
timestamp: August 03, 2021, 04:56:32 am
content: Hi folks.Question for your resonant converter gurus.Is there any easy way or help tool to design an LLC converter transformer with propper leakage to avoid an external inductor?Most of the design papers tell about it but I cannot find a clear description of how to do it.

author: fourtytwo42
author_rank: Super Contributor
timestamp: August 03, 2021, 05:31:40 am
content: I have never heard of a tool, it's something that comes with experience, wind up a few different designs/layouts & test them.The dual bobbin method is probably the most popular as it provides better repeatability than others (e.g. crap or no interleaving).

author: jonpaul
author_rank: Super Contributor
timestamp: August 03, 2021, 12:21:41 pm
content: Bonjour, leakage inductance is a function of the winding layers, type of wire, bobbin length/radial build.Magnetics Designer software had a calculator after a build was designed.Lowest Llkg is a 1:1 ratio, interleaved P-S, and no interlayer insulation.Highest Llkg is the opposite, and with shapes that separate P and S like UU or UI.Our workflow was: Specify all parameters, ratio, Lpri, Llkg, etc.Choose ferrite shape, bobbin, wire types and guages.design windging sheet, use Magnetics Designer to estimate Llkg and other parametersBuild prototype, measure Llkg, refine design,We usually had it done after just 1-2 iterations.Bon Chance,Jon

author: T3sl4co1l
author_rank: Super Contributor
timestamp: August 03, 2021, 01:34:49 pm
content: For the case when windings are tightly coupled, specifically for single layers 1:1, the geometry is close to a parallel-wire transmission line, and that gives leakage inductance proportional to winding length and Zo. It's a bit tougher when sections contain multiple layers (the layers act against each other in this way, reducing the coupling of more distant layers and multiplying the effect of wire length). This is easy to calculate.Things become tricky when the geometry has no close contact, e.g. split bobbin, or windings on separate legs of the core. In this case, it may be easier to work with a coupled-inductors model, and later convert the coupling factor to leakage as needed.Note that the secondary-shorted inductance has a maximum value of the primary magnetizing inductance, when k = 0. That is, there's no difference between Lp(s/oc) and Lp(s/sc). We can't really use an L network equivalent (i.e., series LL, parallel Lm (m = magnetizing), ideal transformer) to describe this situation. We need to use a pi equivalent (with Lp (shunt), Lm (series, m = mutual), Ls (shunt) and an ideal transformer), where Lm can take negative values if we set the transformer ratio to 1. (There always exists a pi or tee equivalent network for any transformer, but the values need not be physical, specifically for the case of large turns ratios and k I think.)So, understand that when k << 1, it may be less meaningful to consider leakage inductance.For a split-bobbin configuration, we can model each winding as a source of MMF (magnetomotive force), and estimate the reluctance between and around them. For an EE core for example, we can model the winding window (which is now partially unoccupied, i.e. the gap between windings) as a wide air gap, and thus it has a relatively high reluctance which shunts some flux from between windings (i.e., leakage), while most of the flux proceeds around the magnetic (core) loop. Note that the core is gapped, so has notable reluctance by itself, and this makes the window leakage relevant. How much so? Depends on the ratio of reluctances. We can adjust the air gap to trade off between them (while adjusting Np, Ns to maintain Lp, Ls as needed).So, say we take the winding window past the middle of the winding, to the end of the core. This has a width of (window length)/2 (assuming the coil is centered), height of (core height or thickness), and length of (window height). This rectangular volume has an inductance of A_L = 2 mu_0 w h / l. (2 because the E core is symmetrical. Or put another way: remember to add up the "height" from both windows either side of the center leg.)FWIW, A "tall" aspect ratio 'E' core, by itself (no E or I completing its magnetic path), gives a mu_eff around 10 I think, so this gives your worst-case A_L (at k = 0). That's with the coil in the middle of the core piece's winding window, on average; it should be somewhat higher if piled closer to the root, and lower close to the open face.We can also add magnetic shunts, if we want lower k while maintaining high Lp and Ls. The shunt width and height should match the core leg's dimensions (to avoid saturation / core heating), and its length (which is the dimension spanning across the winding window) is adjusted to set the air gap for the LL as desired. In the extreme case, the shunt gap is zero and a complete (figure-8) path is made around each coil; now it almost doesn't matter the gap between cores, as this is identical to the case of two complete EI cores butted together. Which is obviously going to be pretty low k. So, between adding shunts (if needed), and adjusting the gap between E's, we can design Lm and LL (or k) as needed.The effect of shunts is to make the winding window air gap shorter and more easily controlled (the more compact air gap has less fringing).As for quantities -- typical gaps will be on the order of 10-30% of core thickness, I think? I don't think there's going to be any easy, fast rule to compute this, and your best luck will be with the winding-window-as-air-gap, and use of gapped magnetic shunts. These can be approximated as rectangular volumes of air gap, and the result should be in the right ballpark, probably uh, better than 50% error, maybe not much better than 20% unless you get lucky? The general case is tough, because fringing fields are everywhere, with overlapping air gaps with fields going diagonally across them, for which rectangular volumes will not give good results. FEA will give excellent results, but of course you need a full 3D magnetostatic simulator for that.The other way to construct these from standard parts, I think is to shim apart two E's with ferrite blocks in the outer limbs, leaving an exaggerated air gap in the center. Use half-bobbins on each E (I guess you could cut down a pair of split bobbins, or use tall "EI" style E's, with two normal bobbins back to back, as a gapped EE assembly), or of course whatever hardware is appropriate (the gapped split bobbins made for this type of transformer). It's probably easier to make and measure an assembly, than to set up a model and simulate it...Have fun!Tim

author: sandalcandal
author_rank: Unknown
timestamp: August 03, 2021, 03:18:56 pm
content: There are some professional and very expensive tools which attempt to enable calculation using FEA simulation. I've also trialled a tool which used machine learning models instead of traditional analytic or simulation methods to estimate transformer parameters. These tools can help you get in the ball park but ultimately it seems trial and error is still required to an extent.I also suggest you consider whether or not you actually want to have your resonance inductance achieved using primary-secondary leakage inductance. This leakage can tend towards causing higher losses both in copper and in core whilst at the same time affecting the LLC dynamics. The leakage inductance will be seen as series inductance on both the primary and secondary. If a discrete/integrated inductor (I've seen a few designs using an additional E stuck to the main E pair like EE∃example integrated inductor patent) then that resonant inductance is only on the primary with the resonant capacitor as opposed to both sides which changes circuit behaviour, particularly if you plan on using synchronous rectification. This white paper here shows some of the effects of using a separate inductor and minimising leakage vs using leakage inductance for resonant inductancehttps://www.transphormusa.com/en/document/design-guide-12v-1200w-high-frequency-llc-converter-design-using-gan-fets/Trying to design (and wind) a transform to have the correct leakage inductance is also a pain in general.Using leakage inductance for the resonant inductance can potentially lower overall cost if you have the capacity to accommodate leakage inductance in your design. Most high performance stuff I see tries to avoid it and use discrete inductance however.

author: Just_another_Dave
author_rank: Regular Contributor
timestamp: August 03, 2021, 06:10:18 pm
content: ANSYS Electromagnetics Desktop includes a tool for designing transformer automatically called PExprt. However, it just supports standard cores.I know people that use it, but I prefer optimizing transformers manually. Nevertheless, I recognize that it is a useful tool for designing transformers quickly

author: Miyuki
author_rank: Frequent Contributor
timestamp: August 04, 2021, 06:20:51 am
content: Quote from: Just_another_Dave on August 03, 2021, 06:10:18 pmCores for hard switching and flyback converters are simple and plenty of tools are available to suggest the size and so.

author: Just_another_Dave
author_rank: Regular Contributor
timestamp: August 04, 2021, 07:45:51 am
content: Quote from: Miyuki on August 04, 2021, 06:20:51 amSorry, I missed the part of your message where you specified that the tool needed to be able to adjust the leakage inductance. In that case I don’t know any tools able to do that. Leakage inductances can change significantly according to the shape of the winding output and, therefore, they are difficult to estimate without a simulation of a detailed model if a low value is used. That’s usually a problem for optimizers as those simulations can be particularly slow

author: Terry Bites
author_rank: Super Contributor
timestamp: August 04, 2021, 09:54:52 am
content: Try Phillips, oops NXP:https://www.nxp.com/products/power-management/ac-dc-solutions/ac-dc-controllers/llc-resonant-converter-design-tool:LLC-RESONANT-CONVERTER-DST

author: T3sl4co1l
author_rank: Super Contributor
timestamp: August 04, 2021, 02:19:07 pm
content: Quote from: Miyuki on August 04, 2021, 06:20:51 amOh and needless to say, those calculations still apply, for the magnetizing inductance and overall core size; the leakage is the tough part. (The waveform of course being more rounded than square so one must use a proper measure of its flux, but flux is flux.)(I think you already know this, just making the note for completeness.)Tim

author: kreutz
author_rank: Contributor
timestamp: August 06, 2021, 06:30:29 pm
content: Check the following link (ACHIEVING THE DESIRED TRANSFORMER LEAKAGE INDUCTANCE NECESSARY IN DC-DC CONVERTERS FOR ENERGY STORAGE APPLICATIONS.pdf )https://www.google.com/url?sa=t&rct=j&q=&esrc=s&source=web&cd=&ved=2ahUKEwjYvKeD_5zyAhU4QzABHZCpBtEQFnoECBQQAw&url=https%3A%2F%2Fnottingham-repository.worktribe.com%2FOutputFile%2F709642&usg=AOvVaw0GhSE_2cmKKXNUrzU9VxdQIt won't let you design for an specific leakage, but at least will let you adjust the leakage inductance a bit

author: TimNJ
author_rank: Super Contributor
timestamp: August 06, 2021, 08:01:44 pm
content: Here are some methods, varying in complexity (and age):https://www.hpe.ee.ethz.ch/uploads/tx_ethpublications/FINAL_PAPER_SUPPLEMENTARY_Schlesinger_TPEL_01.pdfThe Dowell and Rogowski methods are simplest. Honestly, I feel ball-parking it with these types of simple methods is usually...good enough? Will maybe get you within 25%. Make a transformer, check if it matches the model. If not, you can probably add some correction coefficient to your model.Here's Dowell's paper from 1966:http://www.cpdee.ufmg.br/~troliveira/docs/aulas/fontes/05247417.pdfAttached from McLyman, specifically for a split bobbin design.