MC34063 high voltage dc-dc boost converter - Page 1
Forum section: electronics / beginners

author: dazz
author_rank: Frequent Contributor
timestamp: December 30, 2019, 03:41:21 pm
content: Greetings everyone. I've been mulling over the idea of building a dc-dc boost converter that outputs some 300V to power tube amps. I reckon it should be easy to build one capable of powering a two 12AX7 preamp, but I wonder if I can build something with enough juice for an EL84 single ended amp, or even a push pull with two EL84's. This would be used for the plate and screen voltages, I would deal with the heaters separately.I've already done some research and found a few chips that might do the trick, like the MT3608, the LM3478 or the MAX1771, but because I already have some MC34063 lying around, I decided to try this ratio extender circuit:https://www.changpuak.ch/electronics/High_Voltage_Power_Supply_MC34063.phpSo far I have only simulated the circuit, and to my surprise, the simulation throws very promising results. (See attachment 1 for schematic)The guy on the above link who built those boards to power nixie tubes got pretty terrible efficiency figures at 30-40% with this circuit, and the simulation agrees, but then I swapped the switching transistor (FJN13009) for a mosfet (IRF740) and the efficiency (again, in LTSpice) skyrocketed to a 85%. It also seems to make R7 unnecessary, I guess because the mosfet's gate current is negligible so it's not loading pin 2 of the IC, but is this a result that makes sense? Does the mosfet do a better job at staying away from the linear zone hence dissipating less power? Or is this simply an artifact of the simulation?Perhaps even more impressive, or at least it was surprising to me, is that the simulation is perfectly fine outputing 200mA at 300V for 60W at 85% efficiency. According to my research, the MC34063 would exceed it's 1.5A peak current limit at some 50mA in the output at 300V, so 200mA seems way above spec. I will know for sure once I build it, but I thought I would ask in case some of the local experts can help me understand all this stuff.I've read that the MC34063 is old and doesn't support PWM. I guess it works at a fixed frequency and it switches the pulsing signal on and off, and that makes it a worse option than a proper PWM chip, but I don't understand why that may be a problem. Is it because it generates noise as the switching signal turns on and off? I have added a capacitance multiplier to filter out most of the ripple and it seems to work great in the sim, with only 15mV peak-to-peak ripple at 300V. That ripple seems to be at the low frequency (200Hz) at which the mc34063 is controlling the switching signal. The switching frequency itself, at some 20KHz, is completely gone after the capacitance multiplierI'll be using a toroidal inductor for this build, it's [ulr="https://www.ebay.com/itm/3A-6A-Toroid-inductor-Coil-LM2596-Magnetic-Ring-Inductance-22UH-to-100UH/223484117366?ssPageName=STRK%3AMEBIDX%3AIT&var=522239495543&_trksid=p2060353.m2749.l2649"]one of these 3A 470uH coilshttps://www.digikey.es/product-detail/en/aavid-thermal-division-of-boyd-corporation/513102B02500G/HS346-ND/1216353.]. Does that seem adequate?And this heatsink for the mosfet, which according to the simulation, draws 8W with a 200mA load at the output: https://www.digikey.es/product-detail/en/aavid-thermal-division-of-boyd-corporation/513102B02500G/HS346-ND/1216353.It's 11ºC/W so might be a bit too small, not sure.Thoughts? Ideas? Criticism?Thanks in advance

author: T3sl4co1l
author_rank: Super Contributor
timestamp: December 30, 2019, 03:58:26 pm
content: *Cracks knuckles*https://imgur.com/gallery/M1S0DbIEnjoyBut yeah, what you're really looking for is at least UC3843, probably a current transformer to improve efficiency, and definitely a transformer or autoformer so you aren't beating the shit out of the switching parts. Any transformer from a power supply of the same size will do -- they work just fine in forward or reverse. A 120/240V input, 12V output, 50W supply say, will easily get you 300V at 50W from 12V supply.You should have no need of active filtering on a properly designed switcher. Passive filters are simple and effective. The 34063's hysteretic behavior makes a huge mess.Tim

author: dazz
author_rank: Frequent Contributor
timestamp: December 30, 2019, 04:18:17 pm
content: Quote from: T3sl4co1l on December 30, 2019, 03:58:26 pmThanks, Tim. I have a few things to google now, because I don't understand some of the things you say there.I think it's time to crack open The Art of Electronics, chapter 9 and learn how these buggers work before I go on with the build. I should've done that already.I was hoping to get away with not using a transformer, but if that's the right way and I won't need something specific, I'll do it

author: dazz
author_rank: Frequent Contributor
timestamp: December 30, 2019, 04:39:25 pm
content: Quote from: T3sl4co1l on December 30, 2019, 03:58:26 pmWow, I simulated the same circuit at a lower output voltage, higher output current and the efficiency got up to 95% with much less power consumption in the mosfet, so I think I see what you mean now about using a transformer.

author: T3sl4co1l
author_rank: Super Contributor
timestamp: December 30, 2019, 07:36:34 pm
content: Yep. You don't usually see SMPS analysis in terms of reactance, but it generalizes just fine. By driving a higher voltage ratio, or higher current ripple ratio, you draw proportionally more reactive power from the inductor, and that power flows into the transistor and out the diode (or however you like to assign the signs, it's AC).So going for a say 10:1 ratio at 50W might take 10 times more reactive power than 1:1 (flyback type), and if that's at 100% ripple (DCM or BCM), then that's about 500VA reactive power, and you better have one hell of an inductor to handle that with low dissipation (Q >> 100?). And the transistor and diode need to be sized for 500VA despite that you only get 50W from it.Or more generally, for a given factor of range, you need not only components that handle the extra range, butto maintain efficiencyyou needfactor squaredbetter components. And that's just conduction loss alone.So having a modest range, pays. A lot.By using a transformer, you get all the ratio with none of the wasted capacity. You get an additional wrinkle of leakage inductance (and also stray capacitance, at high voltages), but this isn't new as such, it was always present (stray inductance between the transistor, diode and cap); it's only a matter of correct transformer design, and snubbing if applicable.Some more study highlights: current mode control. Draw the switching waveforms (current ramps and voltage flats). Commutation and 1st order parasitics if you like. Derive the transfer function (voltage ratio in terms of D) using the inductor definition. You get the latter non-consecutive two here,http://schmidt-walter-schaltnetzteile.de/smps_e/smps_e.html(i.e. not commutation), but I do suggest understanding how it works, before getting too accustomed to it.Tim

author: dazz
author_rank: Frequent Contributor
timestamp: December 30, 2019, 10:50:27 pm
content: Quote from: T3sl4co1l on December 30, 2019, 07:36:34 pmGreat stuff, as always, Tim. Thank you. I sure want to learn how this stuff works. I'm still a long way to go, but it should be lots of fun.EDIT: And if I could somehow source the right transformer, I guess I could add a 6.3V - 12.6V output for the filaments too, which would be fantastic.

author: David Hess
author_rank: Super Contributor
timestamp: December 31, 2019, 02:11:01 am
content: Quote from: dazz on December 30, 2019, 03:41:21 pmThe MC34063 is a "constant off-time" PWM switching regulator. This means that switching frequency is variable which can be a problem in some applications however it has the advantage of not requiring frequency compensation.Most problems people have with the MC34063 is interfacing its bipolar output to external devices.

author: MagicSmoker
author_rank: Super Contributor
timestamp: December 31, 2019, 12:28:02 pm
content: T3sl4co1l is somewhat infamous around these parts for his dislike of the '34063/33063/3063/etc., but they are enormously popular for several reasons: no frequency compensation required, easy to get working (particularly the buck, as no high-side driver is needed), and the hysteretic control scheme used has the best possible transient performance of any (no Nyquist limit here). All in all, very beginner-friendly and the major downsides like the relatively high voltage drop across the (non-boostrapped) internal bipolar switch and the variable switching frequency being more difficult to filter for EMI are the sorts of things beginners don't usually care about; once you gain some experience - and confidence, from not having something blow up as soon as power is applied, as is so often the case with switchers - then you can move on to current mode controllers driving external switches, etc.However, you won't be getting 300W out of this chip... In fact, even 30W would be a stretch, given that the absolute max ratings are 40V and 1.5A peak, so even if those were continuous/average ratings the best you could hope for is 60W. These ICs - and any with an internal switch, really - also aren't the best choice for transformer isolated converters, but since adding a transformer to the mix makes the design much more difficult this also isn't really something that I'd recommend a beginner at SMPS design attempt. Make a few non-isolated converters - buck, boost and flyback (or buck-boost) - first, then look into how adding a transformer can vastly improve things compared to, say, trying to get 300V out of a boost supplied by 12V (which is totally impractical, even if theoretically possible).

author: dazz
author_rank: Frequent Contributor
timestamp: December 31, 2019, 01:04:29 pm
content: Thanks guys. So I guess the simulation is giving me deceptive results in this case if the MC34063 won't do 60W. I have built a couple boost converters with this same chip, and one buck converter, all of them based on the circuits in the datasheet, then filtered the hell out of them with LC filters, and they worked ok at low current.Anyway, I need to do a lot of reading on the subject and I'll also try building the circuit in the OP to see what it can do and get some practical results.Then I'll look into adding a transformer. Found this one that looks like it should do the trick, right?https://www.ebay.com/itm/12V-To-255V-300V-EC42-EC4045-Horizontal-High-Frequency-Transformer-Inverter/262721601791?hash=item3d2b6d80ff:g:rZwAAOSwYVpdpdSbSeems a bit overkill at 350W though.

author: dazz
author_rank: Frequent Contributor
timestamp: December 31, 2019, 08:05:44 pm
content: Quote from: MagicSmoker on December 31, 2019, 12:28:02 pmCan I ask you why is that the case? Aren't this IC's meant to control the switching? Or do you mean IC's with an internal switching mosfet? Sorry, I'm afraid I'm missing something here

author: MagicSmoker
author_rank: Super Contributor
timestamp: December 31, 2019, 08:15:18 pm
content: Quote from: dazz on December 31, 2019, 08:05:44 pmI was referring to regulator ICs - that is, a controller + internal switch - vs. controller ICs, which require an external switch. Regulator ICs are usually designed for non-isolated applications and, as a result, often only have a single pin for feedback. Controller ICs - which control an external switch, hence the name - typically bring out one or both error amplifier inputs as well as the output which makes it easier to design a compensation network and use an optocoupler for isolating the feedback pathway.Another possible issue with regulator ICs is that is that any ringing or voltage overshoot from the transformer leakage inductance can disrupt the controller's operation (or even destroy it) by causing substrate currents to flow. Basically, once you need to handle more than 10-20W or so you should strongly consider using an external switch.

author: dazz
author_rank: Frequent Contributor
timestamp: December 31, 2019, 08:28:51 pm
content: OK, thanks. Frequency compensation is on my list of things to investigate. In a quick google search I learned it's about avoiding positive feedback/oscillations.Just to be clear, when you say I need an external switch, do you mean the mosfet? I guess that's not it

author: dazz
author_rank: Frequent Contributor
timestamp: December 31, 2019, 08:45:46 pm
content: Oh, OK. I think I see what you mean now. The internal switches are not necessarily mosfets. So for example in the MC34063 the internal switch is that darlington pair at the SWC & SWE outputs, right?

author: MagicSmoker
author_rank: Super Contributor
timestamp: December 31, 2019, 09:10:32 pm
content: Quote from: dazz on December 31, 2019, 08:45:46 pmRight, the switch in a switching power supply can be a MOSFET, BJT, SCR... heck, even a relay (see the "vibrator" type power supply used in early car radios). In this case, you are mostly correct - the actual switch in the MC34063 is Q1 while the driver for it is Q2. Note that the collector for driver Q2 is brought out to a separate pin to allow for more flexibility in driving Q1; most of the time you'll just connect the collectors together in a straight Darlington configuration, though.

author: T3sl4co1l
author_rank: Super Contributor
timestamp: December 31, 2019, 09:35:50 pm
content: Some of those old controllers were also useful (to a limited extent) alone, so could be considered regulators as well. 34063 is kinda like that, depending on how much of the output transistors you use. TL494 has 200mA, 40V "uncommitted transistor" outputs, so can potentially deliver a couple watts by itself.Or, the gate drivers often capable of several amperes (if in pulses), but the always-on output rather constrains direct application of them (you can't very well run a boost converter when there's a transistor firmly clamping the pin to VCC).Tim

author: dazz
author_rank: Frequent Contributor
timestamp: December 31, 2019, 09:50:17 pm
content: Quote from: MagicSmoker on December 31, 2019, 09:10:32 pmDoh! True, it's not a Darlington pair. And the fact that they're BJT's is an issue because it has higher output impedance and loads what's connected to it more than say, fets? Is that right?Also, I just tried TI's Webench and performed a parametrical search. Apparently the boost topology is the simplest one (obviously), push-pull is the most efficient and flyback is a compromise between the two... well I really need to read the relevant chapter in The Art of Electronics to understand how those topologies work.But one of the suggested IC's in webench looks like it could be a good option for my next design: the UC3844A. It's also one of the ones that Tim recomends in that imgur post. It's cheap, through hole and readily available

author: dazz
author_rank: Frequent Contributor
timestamp: December 31, 2019, 09:57:13 pm
content: But I can't seem to find a spice model for that UC3844A... dunno, maybe I've become too reliant on the simulation

author: T3sl4co1l
author_rank: Super Contributor
timestamp: January 01, 2020, 12:38:51 am
content: https://forums.ni.com/t5/Multisim-and-Ultiboard/model-for-UC2842-or-UC3842/td-p/524298?profile.language=enDon't worry about 44, it's the half duty cycle version, aimed at forward converters.Do however use the 43; set the UVLO thresholds to say 5 and 6V in that case.Code:vlow vlow1 0 dc 5vhigh vhigh1 0 dc 5.5Tim

author: dazz
author_rank: Frequent Contributor
timestamp: January 01, 2020, 12:41:17 am
content: Quote from: T3sl4co1l on January 01, 2020, 12:38:51 amDude, you're awesome. Thanks so much

author: dazz
author_rank: Frequent Contributor
timestamp: January 01, 2020, 12:14:33 pm
content: I'm having a look at theUC3843 datasheet. Apparently it's maximum supply voltage is 30V and the maximum output current is 1A. Does that mean this IC can't do more than 30W output?

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 01, 2020, 12:28:11 pm
content: Quote from: dazz on January 01, 2020, 12:14:33 pmNo, that (very rough estimate) only applies to *internal switch*regulatorICs, not *external switch*controllerICs. Note the distinctive terminology used, which applies when searching for parts on the usual distributors, btw.

author: dazz
author_rank: Frequent Contributor
timestamp: January 01, 2020, 12:45:14 pm
content: Quote from: MagicSmoker on January 01, 2020, 12:28:11 pmI see, thanks MS. Hopefully one day I'll be capable of building a full blown AC-DC SMPS with this chip, as I reckon that would provide the best efficiency. Baby steps though

author: dazz
author_rank: Frequent Contributor
timestamp: January 01, 2020, 04:29:35 pm
content: I'm a bit baffled by this. I'm simulating the flyback version of the MC34063 converter at the moment, and surprisingly it seems to be slightly less efficient (82%)I swapped the mosfet for a IRFZ44N with Id=49A & Vds=55V that I think should be suitable for this application, but it's dissipating a whopping 10W and killing the efficiency of the converter. Is it spending too much time in the linear zone or something? (attached screenshot)

author: JackJones
author_rank: Regular Contributor
timestamp: January 01, 2020, 04:35:54 pm
content: Try adding active pull down on the gate. For example see the third image on this page:https://threeneurons.wordpress.com/nixie-power-supply/

author: dazz
author_rank: Frequent Contributor
timestamp: January 01, 2020, 04:48:10 pm
content: Quote from: JackJones on January 01, 2020, 04:35:54 pmGreat, thanks Jack. That's a really cool trick to speed up the switching of the mosfet. According to the datasheet the IRFZ44N has a fall time of 45ns, almost twice as much as the IRF740 I was using in the boost configuration, so that might explain the increased power dissipation in the mosfet, correct?

author: magic
author_rank: Super Contributor
timestamp: January 01, 2020, 05:00:56 pm
content: Re-run the simulation and see if the predicted fall time really is in tens of nanoseconds.My guess is that it'sseveral millisecondsa microsecordBecause it should coincide with the flat part in the middle of gate waveform falling edge.You really need that turn-off speedup before you can start to worry about FET specs. And even then there is nothing to worry about at such low frequencies. 100ns repeated at a few tens kHz is only a few ms total per each second.

author: T3sl4co1l
author_rank: Super Contributor
timestamp: January 01, 2020, 05:18:43 pm
content: Datasheet rise/fall times aren't so much intrinsic to the device, they're typical of the drive conditions. You'll see a lot of Fairchild datasheets with R_G = 25 or 50 ohms, even big transistors for some reason, which of course get laughable times like 400ns. Typically you use a gate driver IC with an output resistance of a few ohms, and the rise time is in the 10s of ns.Regarding the sim, the inductance looks awfully large. Consider how much dI is obtained in the, what, 4us or so expected on-time, at 24V and L. dI = V dt / LTim

author: dazz
author_rank: Frequent Contributor
timestamp: January 01, 2020, 05:25:33 pm
content: Quote from: magic on January 01, 2020, 05:00:56 pmExactly right. 1.15us until it reaches the flat portion of the falling edge. What would be the explanation for this, please? Shouldn't it be closer to the 45ns that the datasheet indicates? I don't know, but looking at the wave form of the gate signal, it doesn't look to me like the time spent in the linear zone is negligible at all, if that's what you meant. What am I missing?

author: dazz
author_rank: Frequent Contributor
timestamp: January 01, 2020, 05:29:26 pm
content: Quote from: T3sl4co1l on January 01, 2020, 05:18:43 pmOh, right. I totally forgot about the transformer inductances. LOLThe thing is that I've been foolishly trying to get the simulation working properly with the old 400V 10A mosfet. So I started trying stupid things like increasing the inductance of the transformer until I finally realised I needed another mosfet... and then I totally forgot about the transformerI'll fix that now.

author: T3sl4co1l
author_rank: Super Contributor
timestamp: January 01, 2020, 05:40:17 pm
content: Quote from: dazz on January 01, 2020, 05:25:33 pmWhat's the test condition in the datasheet? R_G or R_S or something.What is the resistance here?What is the gate capacitance? (Why is that the wrong question to ask?)What is the gate charge? Gate capacitance equivalent?Tim

author: magic
author_rank: Super Contributor
timestamp: January 01, 2020, 05:47:17 pm
content: The flat part is called "Miller plateau". It occurs due to so-called Miller capacitance, which is capacitance from gate to drain. These parts are close to each other and they form a capacitor hidden inside the FET. It's also called "reverse transfer capacitance" and as such you will find it on the datasheet.As drain voltage raises, Cgd charges up and sends current into the gate. The rate of drain voltage raise is limited by the ability of your 200Ω resistor to pull that current out of the gate. In fact, you will see a perfectly linear ramp when you look at drain voltage over time.

author: dazz
author_rank: Frequent Contributor
timestamp: January 01, 2020, 05:53:21 pm
content: So if I got it right, it's some sort of stray capacitance opposing voltage change, hence slowing down the falling edge of the signal. I've read the pcb design is critical in switching regulators and power supplies, I guess stray capacitance is one of the main concerns for these reasons?

author: dazz
author_rank: Frequent Contributor
timestamp: January 01, 2020, 06:02:40 pm
content: Quote from: T3sl4co1l on January 01, 2020, 05:18:43 pmSomething's amiss here. I used the calculator for a flyback converter that you suggested yesterday:http://schmidt-walter-schaltnetzteile.de/smps_e/spw_smps_e.htmlWith 24V input, 300V and 200mA output, the calculator suggests 120uH in the primary and a turn ratio of 80... but as far as I know that would mean 80^2 = 6400 times more inductance in the secondary. That would be 7.68H in the secondary... can't be right. I must be making one of my stupid mistakes

author: T3sl4co1l
author_rank: Super Contributor
timestamp: January 02, 2020, 02:16:24 am
content: The suggested values are for, I think, ripple fraction 10 or 20%. You can always set it lower, and in this case you need to, with a peak current mode controller which will not behave well with a high inductance.You misread the scientific notation: it's 79.81e-3 or 0.07981. It's also labeled N1/N2 so your misread would imply the secondary is 80 times shorter!Tim

author: dazz
author_rank: Frequent Contributor
timestamp: January 02, 2020, 02:56:53 am
content: Quote from: T3sl4co1l on January 02, 2020, 02:16:24 amRipple fraction.. just another term that goes down on my list of things I need to research. Transformers are a world of their own apparently. Jeez, why does this stuff have to be so complicated?Quote from: T3sl4co1l on January 02, 2020, 02:16:24 amDamn it! That was a clear case of confirmation bias. I was expecting a ratio > 1 so I missed the exponential termBut I'm not sure I understand what you mean. If N1/N2 = Nprimary/Nsecondary = 79.81e-3, then the N2/N1 = 0.07981^-1=12.53So if the suggested primary inductance is 120uH, the secondary would be (12.53^2)= 157 times that, hence 120uH x 157 = 18.8mH... assuming I'm not having another brain fart, that is

author: T3sl4co1l
author_rank: Super Contributor
timestamp: January 02, 2020, 04:55:07 am
content: Correct!Tim

author: dazz
author_rank: Frequent Contributor
timestamp: January 02, 2020, 01:35:28 pm
content: Quote from: T3sl4co1l on January 02, 2020, 04:55:07 amBah, I just noticed I misread you too. Anyway, all clear now.I'm getting some mixed results in the simulation now. Interestingly, the active pull down on the gate that Jack suggested worked great in the boost (no transformer) version, boosting the efficiency to a very nice 92%. OTOH I just can't seem to get the flyback version anywhere near as efficient. The active pull down doesn't work unless I reduce the current limiting resistor, which in turn increases the power dissipation in the mosfet, and the efficiency tanks as a result. Just to be sure, does theIRFZ44Nlook like a proper choice for this application?

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 02, 2020, 01:46:49 pm
content: Quote from: dazz on January 02, 2020, 01:35:28 pmThe IRFZ44N is a very old MOSFET so, no, not a good choice for anything these days.The isolated flyback is somewhat tricky to design. Post your LTSpice .asc file here if you want some help with tuning it, but as I already warned, SMPS transformer design is an art unto itself, so expect a lot of pain and escaping magic smoke.

author: dazz
author_rank: Frequent Contributor
timestamp: January 02, 2020, 02:00:28 pm
content: Quote from: MagicSmoker on January 02, 2020, 01:46:49 pmYeah, I'm not going to build the flyback smps anytime soon. I'll make sure I do proper research before I do that.The asc is attached, thanks

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 02, 2020, 03:13:39 pm
content: Quote from: dazz on January 01, 2020, 06:02:40 pmHey, that website is one I often recommend to SMPS beginners but it is giving me a "bad request" error on most (but not all) of the sections. Is it just me, or is it broken for others?Quote from: dazz on January 02, 2020, 02:00:28 pmRight, a bunch of things wrong just at first glance. The transformer phasing is wrong - for a flyback the dot ends are inverted w/r/t each other - and you don't include the H when specifying the inductance, so the primary should just be 120u, not 120uH (you did specify the secondary correctly).Next big issue is there is no RCD or Zener clamp across the primary; switch destruction is guaranteed in the real world, but LTSpice apparently doesn't care...You also need to specify a diode on the secondary with an appropriate voltage rating; a 40V or whatever Schottky ain't gonna cut it with a 300V output.The pot isn't a standard LTSpice component and simulating 3 seconds is about 2.99 seconds too much (quite literally: 10ms should be enough here). I like to set the simulation time to capture 100 to 1000 switching cycles, or 4-10 AC mains cycles, whichever applies.Probably some other things but this should be enough to make your brain hurt for awhile.

author: dazz
author_rank: Frequent Contributor
timestamp: January 02, 2020, 03:19:25 pm
content: Damn, thanks for that, MagicSmoker. I'm hopelessly clueless,Andhttp://schmidt-walter-schaltnetzteile.de/seems to work just fine for me. I tried a few pages and got no errors

author: dazz
author_rank: Frequent Contributor
timestamp: January 02, 2020, 03:24:43 pm
content: Quote from: MagicSmoker on January 02, 2020, 03:13:39 pmAnd sorry about that, forgot about the pot model. I got it somewhere on the web and tweaked it to my liking, but I think it may slow down simulation runs a bit too much, not sure.I can post the model if neededETA: The reason for the long simulation time is because with large filter cap values it takes a while to reach 300V in the output, not sure if there's anything I can do to speed it up

author: dazz
author_rank: Frequent Contributor
timestamp: January 02, 2020, 04:36:15 pm
content: Now we're talking. Fixed the tranny's polarity and increased the gate pull down resistor and got a 96% efficiency already. I still need to find a fast switching diode that will take 500V plus peak reverse voltage and perhaps a better mosfet.ETA: OK, I believe a UF4007 should do for the diode

author: dazz
author_rank: Frequent Contributor
timestamp: January 03, 2020, 03:28:40 am
content: Quote from: T3sl4co1l on December 30, 2019, 07:36:34 pmQuote from: MagicSmoker on January 02, 2020, 03:13:39 pmLearning about this stuff right now. Already understand what you mean by leakage inductance and why I need a snubber network, either RCD or Zener, and I'm in the process of learning how to pick the right clamping voltage to have the best compromise between limiting the voltage overshoot in the mosfet and the power losses in the snubber itself. I can see in LTS the massive spikes in the mosfet's drain voltage (and current). Let's fix that

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 03, 2020, 11:14:23 am
content: Quote from: dazz on January 03, 2020, 03:28:40 amIn the real world - especially if input voltage range is wide - the RCD clamp is usually the better choice (and is more forgiving of sloppy component selection) but the Zener clamp seems to be easier for beginners to understand and is an excellent choice at lower power levels and/or when the input voltage is more or less fixed.The basic idea is to pick a Zener voltage that is around 1.5x to 2x the reflected voltage from the secondary, or the secondary voltage divided by the turns ratio going from secondary to primary. For example, the voltage reflected back to a 1 turn primary from a 300V secondary that has 12 turns is 25V (300 / 12); a Zener (or TVS diode) rated for 36V to 51V (using standard values) would be appropriate. You want the clamping voltage to be as high as possible to both reset the leakage inductance quickly and minimize the loss in the clamp - current will not flow out of the secondary until the leakage inductance is reset - but not so high that you have to increase the voltage rating of the switch so much that efficiency suffers from the inevitably higher Rds (keeping price constant). Note that the Zener clamp doesn't help snub any high frequency ringing - it can make it worse, in fact - so you might need a light RC damper across the primary, too (but this is getting into one of motivations for using an RCD clamp... a bit advanced of a topic for the moment).

author: dazz
author_rank: Frequent Contributor
timestamp: January 03, 2020, 11:40:30 am
content: Quote from: MagicSmoker on January 03, 2020, 11:14:23 amAwesome, thanks. I've been trying zeners in that range of voltage, and sure enough, they don't do anything for the large, almost instant spikes in the rising edge of the waveform.I'm currently trying this:What I don't understand is why the clamp is connected to the positive line instead of GND, wouldn't that derive the spikes right back into the tranny?I will also try the RCD clamp, good to know it's a better option here

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 03, 2020, 12:09:19 pm
content: Quote from: dazz on January 03, 2020, 11:40:30 amYou can return the clamp to ground - ie, wire it across the switch - but then it has to withstand the sum of the input voltage + the reflected voltage (ie - same as the switch). The transformer (and switch) don't really care one way or the other. The same applies to the RC damper, if used. BTW - a good rule of thumb for the RC damper is to make C about 3x the output capacitance of the switch and R somewhere around 1x to 3x the characteristic impedance of the LC network formed between the total capacitance of switch and damper and the leakage inductance. For example, if there is 2.4uH of leakage and the switch output capacitance is 50pF then a damper comprised of 150pF and 100-330R will likely clean up the highest frequency ringing (the lower frequency ringing in the flyback is between the magnetizing inductance and the lumped capacitance and can't really be suppressed as it is invariably too close to the switching frequency).One other thing is that 1% leakage is a more realistic minimum for a typical E-core design (and even that requires considerable care in winding geometry), so set the coupling coefficient, K1 (2, 3, etc.) to 0.995 (leakage factor is 1-K2).

author: T3sl4co1l
author_rank: Super Contributor
timestamp: January 03, 2020, 12:29:15 pm
content: As long as C2 is close to the snubber and transistor, it acts to turn the power rail into a supernode, i.e., in AC terms, equivalent to GND. In an AC-equivalent circuit, all rails are drawn as GND, which looks very strange the first few times you see it, but helps illustrate how things work.Tim

author: dazz
author_rank: Frequent Contributor
timestamp: January 03, 2020, 12:41:29 pm
content: Thanks guys. Plenty of things to google in those two last posts. If you find it takes a while to load google.com, bare with me, this is important

author: David Hess
author_rank: Super Contributor
timestamp: January 03, 2020, 02:31:46 pm
content: Quote from: dazz on January 03, 2020, 11:40:30 amAs is usually the case, the schematic does not reflect what is actually happening with the circuit layout. (1) The leakage inductance spike which needs to be snubbed comes from the transformer primary so minimizing the loop area means placing the snubber electrically close to the primary connections.As T3sl4co1l points out, C2 makes the positive and negative input the same AC ground but besides the issue MagicSmoker identified, this also increases the loop area to extend through C2 lowering the effectiveness of the snubber.(1) There is a way to indicate proper layout on the schematic with 45 degree connections but hardly anybody bothers.

author: dazz
author_rank: Frequent Contributor
timestamp: January 03, 2020, 03:37:11 pm
content: Damn, of course. A cap is a short for AC signals. Electronics 101.EDIT: The zener seems to be clamping the signal, but as anticipated by MagicSmoker, it's not doing a great job.I'll try the RCD snubber next

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 03, 2020, 04:52:27 pm
content: R2 in your circuit is a bit of kludge in that it provides some damping whenever the Zener/TVS conducts, but it has to be a realistic value given the expected peak current because the voltage drop across it adds to what the switch must withstand (keep in mind the leakage inductance acts like a current source).

author: dazz
author_rank: Frequent Contributor
timestamp: January 03, 2020, 07:53:46 pm
content: Quote from: MagicSmoker on January 03, 2020, 04:52:27 pmI've tried anything from 200 up to 10k for R2 and a bunch of different zeners from 15V to 100V with no apparent luck, not sure if I'm doing something wrong, but no combination reduced the power dissipation in the mosfet significantly.So D4 is a TVS? I was wondering what it's doing there. Picked a uF400x because I read somewhere that it needs to be a fast switching diode. But then again the MRA4007 that they're using in the schematic I posted above is a standard recovery diode according to the datasheet...

author: David Hess
author_rank: Super Contributor
timestamp: January 03, 2020, 08:20:48 pm
content: The diode has to turn on fast which is generally not a problem even with a standard recovery diode unless it is defective. Diodes can do weird things sometimes. The same of course applies to the zener or TVS. Since high voltage zener and TVS diodes work in avalanche mode, I would expect them to turn on very quickly in reverse breakdown.The alternative snubber uses a diode in series with a parallel RC load so the capacitor absorbs the spike and the resistor slowly bleeds off charge. No zener or TVS is used.

author: dazz
author_rank: Frequent Contributor
timestamp: January 03, 2020, 09:02:38 pm
content: Quote from: David Hess on January 03, 2020, 08:20:48 pmOK, so you mean only the Ton time matters here, because that's when the spikes happen, right?

author: David Hess
author_rank: Super Contributor
timestamp: January 03, 2020, 10:31:30 pm
content: Quote from: dazz on January 03, 2020, 09:02:38 pmExactly, and even slow diodes can turn on very quickly. Some diodes do have problems turning on quickly under some conditions but I think this is a manufacturing problem. It happens even with fast recovery diodes.

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 03, 2020, 10:49:36 pm
content: Quote from: dazz on January 03, 2020, 07:53:46 pmYep, you're doing something wrong... namely, throwing random values at the schematic to see if anything works. Sometimes you can learn a lot that way, but most of the time you just end up jogging in place (ie - going nowhere).In this case, first look at the peak primary current then pick a resistor value that will result in around 10-20V of drop at that current. For example, if the peak current is 2A then a 5-10R resistor would be more appropriate, not 100R or higher! Keep in mind the Zener/TVS (D5 in the above) diode needs to be able to handle this peak current, too, and it absolutely must have a breakdown voltage significantly higher than the supply voltage (more specifically, it needs to be higher than the reflected voltage from the secondary, as I already explained).

author: dazz
author_rank: Frequent Contributor
timestamp: January 03, 2020, 11:01:33 pm
content: Quote from: David Hess on January 03, 2020, 10:31:30 pmCool, thanks David.BTW, I wasn't getting very good results in the sim with any combination of components in the snubber network, so I went back as I usually do, to re-read the thread. Tim said a few pages back that my tranny inductors were far too large. Trying much lower values now to reduce the leakage inductance, and efficiency/power loss in the mosfet are already much improved. When I find the optimal values of inductance for the transformer I'll go back to testing snubbers in the sim. I know this trial and error strategy is probably not the best way to go about understanding how thing actually work, especially since these are no real world tests, but I hope this will give me a general idea of what works and what doesn't, and then I can better understand why once I google the concepts. I dunno, we'll see.

author: dazz
author_rank: Frequent Contributor
timestamp: January 03, 2020, 11:21:10 pm
content: Quote from: MagicSmoker on January 03, 2020, 10:49:36 pmYeah, you're right, I need to stop trying to brute force things by mindlessly trying values when I don't know what they'll do. And thanks for those pointers. May I ask why the 10-20V drop in the resistor? I would have thought that it would need to sink the same voltage as the zener clamp, or about 2x the reflected voltage in the primary, in my case 2x(300V/12.5)=2x24V=48V

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 03, 2020, 11:41:34 pm
content: Quote from: dazz on January 03, 2020, 11:21:10 pmThere's no hard-and-fast rule for sizing the resistor - in fact, it can be eliminated if an RC damper is also present - so don't get too hung up on the voltage drop; I just ballparked something that looked like it would be about right based on experience.

author: dazz
author_rank: Frequent Contributor
timestamp: January 03, 2020, 11:51:53 pm
content: Quote from: MagicSmoker on January 03, 2020, 11:41:34 pmThe RC dumper is in my things-to-google list, thanks again, MS

author: magic
author_rank: Super Contributor
timestamp: January 04, 2020, 09:41:10 am
content: Actually, I have a dumb questionWhat if we take the RCD capacitor and return it to the negative supply, in parallel with the switch, rather than to the positive supply?This would seem to absorb the energy of the whole loop, including capacitor and PCB, not just the transformer. Energy which otherwise rings with the switch and/or avalanches it, I suppose.The only drawback I see is that now the process of discharging the clamp needs to deal with supply ESL, but that's relatively slow and resistively damped so meh (?)

author: T3sl4co1l
author_rank: Super Contributor
timestamp: January 04, 2020, 11:21:17 am
content: Yes, it is electrically preferable to put the clamp across the switch.It is done to the supply, when the supply inductance is low enough not to mind; the advantage is easier layout.Instead of an RCD clamp snubber, an RCD rate snubber is also often seen, which allows some savings in turn-off losses. The RC can also be dimensioned to dampen the transformer's free ringdown waveform. (This tends to be impractical when leakage is high, as the rate snubber will not significantly reduce the peak voltage.)Tim

author: dazz
author_rank: Frequent Contributor
timestamp: January 04, 2020, 02:20:50 pm
content: I found what looks like a great series of videos about flyback design.https://www.youtube.com/user/rbola35618/search?query=analysis+and+design+of+a+flyback

author: dazz
author_rank: Frequent Contributor
timestamp: January 04, 2020, 11:34:22 pm
content: OK guys, I'm really confused about the results I'm getting with different snubber configurations. I attached screenshots of the drain voltage with no snubber, with a zener clamp, with a RCD snubber, and a combination of zener+RCD. First thing I don't understand is the zener clamp doesn't seem to be clamping at all. The zener in the pic has a 20V reverse voltage, yet most of the signal while the mosfet is on is above 20V. It does seem to do a good job at eliminating the ringing while the during the off time of the mosfet though (pic #2). The RCD filter, OTOH, does the oposite apparently. It filters out the ringing while the mosfet is on (pic 3). So I tried a combination of both, which seems to be working pretty well (pic 4).One thing I noticed is that the voltage in the drain while the mosfet is off, is not 24V like in the input, but 48V. It must have something to do with the transformer, because if I decrease the inductance in the secondary, off drain voltage increases... but shouldn't it be the other way around?

author: T3sl4co1l
author_rank: Super Contributor
timestamp: January 05, 2020, 06:30:23 am
content: 24V zener, 24V supply, what's 24+24V?RCD, try more like 1k and 100nF.Tim

author: magic
author_rank: Super Contributor
timestamp: January 05, 2020, 07:56:38 am
content: The peaks are higher than 24+24V. That's because primary current at turn-off is presumably several amps. This means significant voltage drop across the 10Ω resistor and trouble for the tiny 0.5W zener.BTW, it seems that off-time primary voltage approaches zener breakdown voltage. That's not a good idea, you may end up dissipating all flyback transformer energy in the zener rather than the secondary

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 05, 2020, 01:10:31 pm
content: Quote from: dazz on January 04, 2020, 11:34:22 pmWell, someone's not paying attention. Go back and (re-)read posts #44, 46 and 56 as I already covered all of the above questions!The voltage across the switch when it turns off is the sum of the supply voltage and the "reflected" output voltage (ie - the output voltage transformed by the sec:pri turns ratio [note the order]). If the output is 300V and the pri:sec turns ratio is 1:12, then the ratio going in the other direction is 12:1, so the secondary voltage will be divided by 12 when seen at the primary, or 25V in this example. The reflected 25V from the secondary adds to the 24V supply to give a minimum voltage the switch must withstand at the instant of turn-off of 49V (not including ringing, leakage spikes, etc.). If the Zener clamp across the primary breaks down at a voltage lower than 25V then energy stored in the transformer during the switch on time will preferentially exit through the clamp during the switch off time (ie - the flyback period, hence the name), rather than from the secondary. To use a classic military abbreviation, that's NFG, hence why I suggested in one of the aforementioned posts to set the Zener voltage to between 36V and 51V.

author: dazz
author_rank: Frequent Contributor
timestamp: January 05, 2020, 01:44:45 pm
content: Yes, I need to take a break from the sim and read more instead. The reason I used such a low voltage zener is simply because it seemed to work. The higher voltage ones weren't clamping at all, perhaps because of the high frequency of the ringing, not sure. It's also obviously true that the zener I picked was dissipating far too much power, some 15W on average. I probably should also swap the mosfet for one with lower Cds, I think that will reduce the ringing a bit, correct? Is the output capacitance in the datasheet the same as Cds?ETA: One question, please. Is the snubber supposed to improve overall efficiency? Or it simply lowers the dissipation in the mosfet by the same amount the snubber dissipates?

author: dazz
author_rank: Frequent Contributor
timestamp: January 05, 2020, 03:09:53 pm
content: Quote from: MagicSmoker on January 03, 2020, 12:09:19 pmOK, just reread this and I see now you answered many of my questions there. Yep, I'm a bit slow, sorry about thatETA: This looks much better, right? (Added screenshot)

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 05, 2020, 03:12:28 pm
content: Quote from: dazz on January 05, 2020, 01:44:45 pm1. R2 is artificially boosting the clamping voltage at the instant of turn-off; try setting it to 1m to effectively take it out of the circuit without actually removing it.2. Cds is a bit of a moving target because it varies inversely - and non-linearly - with the applied drain-source voltage (when off, of course; this capacitor is obviously shorted when the MOSFET is on). A MOSFET with lower Cds with all other parameters (except cost) being the same will store less energy which eventually needs to be snubbed so, yes, that is a good direction to go. It's a bit of a 2nd order problem at the moment, though; in other words, concentrate on getting the converter to work properly first.Note, also, that Cds (plus other stray capacitances) forms two different resonant networks in the flyback: with the leakage inductance of the primary when the secondary diode is conducting, and with the magnetizing inductance if the diode turns off some time before the switch turns on again (ie - operating in Discontinuous Mode). Basically, you can damp the former but not the latter.3. Standard dissipative snubbers just move losses from the switch to themselves, they don't reduce losses (they actually increase the total loss somewhat because of pesky thermodynamics). There are a class of so-called "lossless" snubbers that utilize resonant networks to recycle energy from switching and/or ringing back to the supply (or output), but don't even think about messing around with them for now.

author: dazz
author_rank: Frequent Contributor
timestamp: January 05, 2020, 03:32:34 pm
content: Quote from: MagicSmoker on January 05, 2020, 03:12:28 pmOK, I have calculated my leakage inductance at 60nH (which seems low, right? I guess I cheated a bit by lowering the inductances of the transformer too much) and the mosfet output capacitance is 380pF according to the datasheet, I guess I can do a lot better than that. So with the clamping voltage at 50V I still get spikes in excess of 60V, a 80V mosfet would be the way to go, correct?

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 05, 2020, 09:12:04 pm
content: Quote from: dazz on January 05, 2020, 03:32:34 pmGah... It took me less time to just edit your .asc file than type another response. My usual disclaimer applies: I did not optimize everything to leave something for you to do, but I got you into the ballpark.That ratio extender trick isn't needed if you select a reasonable base switching frequency vs. inductance vs. peak current so that the flyback doesn't really need to go much above 50% duty and operates as close to continuous current mode as possible without actually entering it. Hence why the timing capacitor was reduced and the primary inductance increased.An RC damper was added across the switch to deal with the high frequency ringing - it only costs about 0.3W of loss and stops your flyback from doing double duty as an AM (or FM!) radio transmitter. As mentioned earlier, this damper can be placed across the primary, instead, I just felt it kept the schematic neater putting it across the switch.The primary clamp is now an RCD type with the (51V) Zener acting as more of a backstop. Loss in the clamp once the converter reaches equilibrium should be around 0.5W or so with peak voltage well controlled for a 100V MOSFET.Output capacitor increased to a more realistic value and output diode changed to a 1A/600V SiC Schottky, though as long as you keep the flyback in DCM you can use a conventional fast recovery diode (ie - with <100ns reverse recovery time [given as tt in the .model statement, btw]).Finally, the simulation parameters were tweaked to give more accurate results and the total time reduced to just a little more than is needed for the output voltage to reach equilibrium (at which point the '34063 goes into burst mode). All in all, waveforms look pretty realistic now and you have a reasonable jumping off point to keep tinkering.

author: dazz
author_rank: Frequent Contributor
timestamp: January 05, 2020, 09:28:17 pm
content: Quote from: MagicSmoker on January 05, 2020, 09:12:04 pmThanks, much appreciated. I will keep on searching for info to learn how all this works and do those calculations myself, if I can.ETA: WTF? 98% efficiency? I know this is just a simulation, but wow!

author: dazz
author_rank: Frequent Contributor
timestamp: January 23, 2020, 03:39:52 pm
content: I've put aside the flyback research for the moment while I build and test the original boost converter.I have most of the components already, but I've come to the realization that I need to learn more about inductors and how they're built before I do this. The thing is I simply searched for 470uH 3A toroidal inductors in ebay, and gotthese crappy onesthat I'm pretty sure won't even handle 1A. What's worse, the ones I got had 0.3mm wires, not the 0.7mm advertised. So yeah, all but useless.I've spent an afternoon googling about toroidal inductors and it's a bit daunting TBH. Apparently it's not as simple as picking a core size that can fit the necessary number of turns of the necessary wire gauge to obtain the desired inductance and current rating/resistance. Looks like there's a host of different core materials that affect the calculations.I'm going by this right now:https://cromwell-intl.com/radio/copper-wire/There's a table specifying the different inductance indexes for every core type, and the formulas to derive the necessary number of turns for a given target inductance. Seems straightforward enough.I also did a search in Digikey to get an idea of the resistance and core size of inductors in the ballpark of what I need. Most seem to be 1.280" Dia x 0.650" W (32.51mm x 16.51mm). Unfortunately the datasheets I checked don't say anything about core material or wire gauge.Is there a standardized method to derive the kind of inductor you need? Would you star by picking the core material, then the size, wire gauge, etc? The problem is, as far as I can tell, all those things are interconnected: a thicker gauge requires a larger core. Materials with larger Al index can be smaller, but then the wire needs to be thinner too, but a thinner wire will produce more resistance with the same number of turns... What a mess!I'm sure there's an easy way to approach this. Any pointers appreciated

author: T3sl4co1l
author_rank: Super Contributor
timestamp: January 23, 2020, 05:25:56 pm
content: Yes, start with core type, roughly.Typically 20 < μ_r < 60 gives best compromise between core size and copper losses, with values outside of this range being useful in less common conditions (typically, lower mu at higher frequency; or higher mu where number of turns or wire length is weighted more heavily than energy density, HV flyback transformers being a typical example).Typical powdered iron materials are #18 and #52 (Micrometals mix numbers), MPP, or branded formulations like Hi-Flux, Sendust/Kool-Mu, etc.Typical ferrites are gapped shapes. Ferrite toroids are either hi-μ or hi-freq materials, and you can't easily gap a toroid, so if we're talking just toroids, yeah.Core size: we can use an energy argument here to find minimum. We may however still need a larger core, or lower ripple fraction or lower frequency in the design (higher inductance, slower Fpwm, slower control loop) to deal with losses, in which case that needs to be fed back into the design, and the inductor design/selection process starts over.Energy storage is given by the Maxwell stress: σ = B^2 / (2 μ) This has units of pressure, which is also units of energy density. Set B = Bsat, then divide your energy storage requirement E = 0.5 L I^2, by this, to get core volume v_e. Select parts with equal or greater volume.Note this is inversely proportional to μ, so a lower μ core stores more energy -- but it also requires more turns for the same inductance, which increases copper losses. The preferred range of μ is driven by typical frequencies and the resistivity of copper, of all things. (After all, if we had casual room-temperature superconductors, we wouldn't need cores at all, just pile on thousands of turns of hair-fine wire and not care.)Core loss: determine Bpk for your application (i.e., the ripple fraction). Find the material's core loss curves, and match up Bpk and F. The power loss density will typically be 100-1000 mW/cm^3 (or kW/m^3 or however they label it). Higher losses are acceptable in smaller cores (say 500 for T50 size cores?), because the surface area/volume ratio is more favorable, and lower for larger (say 100-200 for T300 size?). You can calculate or find the core surface area, and use a typical convection rate, to solve for temperature rise, and in turn, loss density.(Again, as a density, multiply by v_e to get watts.)Finally, check that your wire fits the core. Typically, Bsat will require 200-2000 At, and with passive convection, over 500-1000 At is where things start getting impractical (in terms of winding area and temp rise).If any of these conditions is not met, try the next larger size core, etc.Should be able to find candidates in a handful of iterations, and this method can also be applied to a table of parts to automatically select or rank them. There are unfortunately too many variables to write out a closed-form solution -- although this is a bit easier when manufacturers provide core loss formulae.Going back to ferrite: choose μ the same way, then calculate the air gap based on μ_avg or μ_eff (which, I don't remember the formula offhand but it's easily found; or use the figure from the datasheet, if pre-gapped cores are used).Tim

author: Siwastaja
author_rank: Super Contributor
timestamp: January 23, 2020, 05:42:22 pm
content: Also note this good source of practical information:http://pigeonsnest.co.uk/stuff/core-saturation.html

author: dazz
author_rank: Frequent Contributor
timestamp: January 23, 2020, 05:45:06 pm
content: Thanks guys, apparently this is even more complex than I thought. I'll take my time to educate myself on the topicETA:Quote from: Siwastaja on January 23, 2020, 05:42:22 pmLoving the rant and the colorful language there, LMFAO.Great resource, bookmarked!

author: dazz
author_rank: Frequent Contributor
timestamp: January 25, 2020, 11:45:58 pm
content: OK, so I finally got around to building the prototype of the boost converter, schematic attached.So far it's a massive failure. I feed it 12V and I get 11.7V at the output, and the mosfet get's rather hot at some 100ºC after a minute or so. That's with no load at the output. It's also drawing 1.6A from the power supply, I'm guessing most of that is being dissipated by the mosfet. I have this heatsink for the mosfet:https://www.ebay.com/itm/2PCS-TO-247-TO-220-34x38x12mm-Aluminum-Heatsink-Cool-Heat-Sink-Sound-Transistors/223287755102.The inductor is a 100uH 0.4 ohms toroidalAny off-the-top-of-the-head ideas on what I should check, please? My equipment is very limited unfortunately. No oscilloscope available for now

author: Siwastaja
author_rank: Super Contributor
timestamp: January 26, 2020, 03:59:30 pm
content: Get an oscillosscope, sorry but you just can't do switch mode converters without one. A cheap Rigol will do fine.

author: dazz
author_rank: Frequent Contributor
timestamp: January 27, 2020, 03:47:56 pm
content: Quote from: Siwastaja on January 26, 2020, 03:59:30 pmYeah, I know, I'll see what I can find. Thanks SiwastajaBTW, guys, I sort of got it working. Sort of.I replaced de 1N5817 diode with a proper rectifier diode and it's now outputting hundreds of volts. How many? Not sure because the hysteresis is insane! With a 100uH toroidal inductor, and the pot set to the minimum output voltage that should produce some 100V, it jumps to 320V, Vref to 3.5V, then the IC apparently turns off and the voltage drops to 100V, Vref to 1.2V, and back again to the massive spike.I then replaced the inductor with a 470uH one, and as a result, I get 460V at the output and 5V at Vref. It stays like that, as if the IC never stops switching. Had to stop the test because the output cap is rated at only 450V. Also the inductor gets scorching hot, which I believe is to be expected if it's conducting all the time.

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 27, 2020, 05:14:15 pm
content: Put a load on the output. Running a boost (or flyback) converter without a load is just asking for trouble.And yeah, get a scope. Heck, even one of those janky little DSO Nano's is better than nothing.

author: dazz
author_rank: Frequent Contributor
timestamp: January 27, 2020, 05:19:54 pm
content: Quote from: MagicSmoker on January 27, 2020, 05:14:15 pmOh, I thought it was only flybacks that needed a load to be stable. Right now I have a tiny 680K load, that's less than 1mA, will try that, thanks.And I'm already shopping around for a scope

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 27, 2020, 06:22:33 pm
content: Quote from: dazz on January 27, 2020, 05:19:54 pmBasically, any switchmode converter in which power is supplied to the load during the switchofftime is going to be unstable without a load, and the boost converter does just that, same as the flyback.

author: dazz
author_rank: Frequent Contributor
timestamp: January 27, 2020, 06:30:14 pm
content: Quote from: MagicSmoker on January 27, 2020, 06:22:33 pmIt's the sudden current spike at the inductor when the magnetic field collapses that does this, right? I noticed the effect got progressively less noticeable as I decreased the inductance of the coil. I'm at 33uH right now.Building a constant current load with a mosfet now to test it under load. What kind of load current should I expect to make it stable? Because at 300V even a few mA will mean a few watts of wasted power and a pretty large dummy load resistor. I guess I'll find out soon enough

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 27, 2020, 08:36:15 pm
content: Quote from: dazz on January 27, 2020, 06:30:14 pmKinda/sorta/almost. The nature of an inductor is to resist a change in current, right? When the switch turns on in a boost (or flyback) converter current builds up (relatively) slowly and linearly, and when the switch turns off the current that was flowing in the inductor wants to keep flowing, creating whatever voltage is necessary to maintain said flow. Basically, when the switch turns on the boost inductor acts like a current sink and when the switch turns off it acts like a current source.Quote from: dazz on January 27, 2020, 06:30:14 pmUsually somewhere between 1% and 10% of full load power is necessary to maintain regulation. I don't know how your ratio-extender kludge with the '34063 is going to behave in real life, however; it certainly didn't behave well in the LTspice simulation...I would also recommend against a beginner using a MOSFET current sink (load) to test switchmode converters because that pits two control loops against each other (the current regulation loop on the constant current load and the voltage regulation loop of the switcher), however given the '34063 is a hysteretic controller there is less risk of inciting full blown oscillations here (with the inevitable release of magic smoke soon thereafter).

author: dazz
author_rank: Frequent Contributor
timestamp: January 27, 2020, 09:52:07 pm
content: Quote from: MagicSmoker on January 27, 2020, 08:36:15 pmWhat's wrong with it, please? I thought it was performing pretty well in the sim with a very respectable (I think) 85% efficiencyQuote from: MagicSmoker on January 27, 2020, 08:36:15 pmThat might explain why I can't seem to get it to work with the boost converter. This is the one I'm trying:

author: MagicSmoker
author_rank: Super Contributor
timestamp: January 27, 2020, 11:24:32 pm
content: Quote from: dazz on January 27, 2020, 09:52:07 pmIt made the '34063 model run in bursts with wildly varying pulse widths from cycle to cycle, some of which were way too long (ie - allowing current to build up way too high in the boost inductor). That's a good way to blow up FETs. That said, this could be the result of a dodgy '34063 model. I didn't investigate the issue too deeply (not at all, actually...).Quote from: dazz on January 27, 2020, 09:52:07 pmThat circuit is almost guaranteed to oscillate because the op-amp is saddled with driving a capacitive load directly (the gate of a MOSFET is, essentially, a capacitor). At the very minimum there needs to be a resistor in between the output of the '358 and the MOSFET gate (take feedback directly from the op-amp output, however); 100R should work fine here. Also use a FET with an appropriate voltage rating for the task (the IRFZ44 is rated for 55-60V max).That said, it's hard to argue with the cheap-n-cheerful approach of just tacking on a bunch of 330k/0.5W metal film resistors in parallel as needed to provide about 1mA of load each at 300V.

author: dazz
author_rank: Frequent Contributor
timestamp: January 28, 2020, 03:25:54 pm
content: Thanks again, MagicSmoker. I'll try that resistor in series with the gate. It's the same principle as a grid stopper, right? I have a proper mosfet for the job there, an IRF740 that can handle up to 400V.I removed the 100uF input cap of the converter and that further improved the stability of the circuit. Under a 11mA load, with 111V at the output through a 20W 10K resistor, it is stable, but it's not working properly: with 12V at the input, the current draw there is 730mA. So that's 1.2W at the output and almost 9W drawn at the input. Horrible. Vref is stuck at 1.34V which makes no sense to me. Looks like the MC34063 never switches off. I believe most of the wasted power is being dissipated at the 50uH inductor I have in there right now, because it gets very hot while the mosfet runs cool as a cucumber.The inductor is a toroidal one, about .77 inches of diameter and AWG 20 wire (or something like that)This thing is starting to look like a lost cause.

author: VEGETA
author_rank: Super Contributor
timestamp: February 01, 2020, 04:40:35 pm
content: I was thinking about making an isolated DC-DC converter to get 6-8.4v from li-ion pack and convert them to 12v DC at say 200mA max. I found this IC MC34063 to be cheaper than LT8300 and I found this little SMD transformer:https://lcsc.com/product-detail/Power-Transformer_TDK-ATB322515-0110-T000_C415291.htmlI wonder if this minimal circuit works with these parts?I don't need fancy stuff, just continuous 12v isolated supply and I want it to be less than read-to-use modules like the 0505 or 1212 ones. These ones will require more circuitry to fit my application such as boost or buck to make the signal suitable for each module.Looking forward to your feedback!Note: I couldn't find an ltspice model for MC34063, where and how can I make it work?thanks!

author: dazz
author_rank: Frequent Contributor
timestamp: February 01, 2020, 04:50:33 pm
content: Quote from: VEGETA on February 01, 2020, 04:40:35 pmhttps://www.eevblog.com/forum/beginners/simulate-mc34063-on-ltspice/?action=dlattach;attach=38508

author: VEGETA
author_rank: Super Contributor
timestamp: February 01, 2020, 04:55:50 pm
content: Quote from: dazz on February 01, 2020, 04:50:33 pmthanksi downloaded a .zip file and now trying to make it do the output voltage i want. I am not successful yet.

author: T3sl4co1l
author_rank: Super Contributor
timestamp: February 01, 2020, 05:52:22 pm
content: On a related subject, hah, just now realizing RECOM makes Nixie driver supplies:https://www.digikey.com/product-detail/en/recom-power/R12-100B/945-2050-5-ND/3775934Affordably priced too, despite RECOM usually being the more expensive brand.Tim