The size miniaturization of a single antenna unit is achieved through an optimized placement of slots and extended arms. The quad-antennas are then placed orthogonally to achieve antenna diversity. The antenna resonates at 3.56 GHz and 5.28 GHz having 2:1 VSWR fractional bandwidth of 1.82% and 2.12%. The proposed resonator provides 88.34% and 79.28% efficiency at lower and upper bands, respectively. The antenna is an exceptional radiator regarding MIMO diversity performance owing to high inter-element isolation. The values of envelope correlation coefficient < 0.05, channel capacity loss is nearly 0.1 bits/sec/Hz, and total active reflection coefficient is − 24.26. The full ground plane profile aids in high directivity and cross-pol isolation. The antenna exhibits a gain of 4.2 dBi and 2.8 dBi, respectively, justifying intended application requirements. There is a good coherence between simulation and experimental results. The self-decoupled antenna poses its application in 5G and WLAN Communication Applications.A four-port MIMO antenna with high isolation is presented. The antenna is primarily envisioned to cover the n48 band of Frequency Range-1 (FR-1) with TDD duplex mode. The engineered antenna has electrical dimensions of 90 × 90 × 1.57 mm Owing to such demand, antenna performance enhancement seeks continuous research and development2. High-speed data rate is a key demand for emerging technologies such as virtual reality, online gaming, and smart city3. The multiple antenna structural designs allow high data processing and throughputs to be available using advanced radio propagation schemes. The 5G new radio (nr) technology, a unified and capable wireless air interface, envisions improved reliability and extremely high data rate4. 5G NR supports adaptive bandwidth support, less time lag and minimal network losses. The carriers can be simultaneously aggregated over the spectrum. With 5G momentum, the communication networks shall provide end-to-end connectivity to user equipment directly connected to the 5G core communication network. Multi-network connectivity shall enable users to achieve the Gigabit per second data rate5. The 4G services do not have the potential to cater for the requirement of data traffic created through enormously demanding applications such as live video streaming, cloud services, and online gaming, to name a few6. The 5G technology is now being implemented by the majority of nations across the globe to accomplish the need to handle large data7. The centimetre and millimetre wave spectrum of 3–300 GHz has been quite effectively utilized for 5G communication technology for achieving extremely high throughput in the order of some Gigabit per second. The additional benefit of targeting this frequency is that a lower spectrum is utilized in parallel for other communication applications such as ISM, Wi-Fi and WiMAX. In contrast, a higher spectrum can be available for 5G wireless communication applications.The antenna is a vital component of the communication system. The data communication requirement continuously evolves about data transmission increase8. The optimal spectrum efficiency and resource sharing in channel allocation enable more end-user devices9. The radio propagation issue of fading is addressed as each antenna faces inconsistent multipath fading and gets nullified. A wide range of multiplexing techniques exist to provide a large degree of freedom in 5G technology10. The MIMO technology significantly reduces radio propagation issues through multiple transmit and receive antennas, multiplexing techniques, and high spectrum efficiency. The MIMO antennas should be able to provide high inter-element isolation between antenna elements and gain13. The high inter-element isolation aids in achieving higher throughput from the MIMO antenna system causing better communication efficiency16. The MIMO antennas can exhibit increased channel capacity and enhanced link reliability through multipath propagation techniques. The MIMO antennas having high inter-element isolation is a key design challenge for antenna designers, especially with a closely packed device environment17. The mutual coupling between the MIMO antenna elements primarily causes E-plane radiation to deteriorate due to surface current flows in the neighbor element. However, H-plane would not get significantly affected due to magnetic coupling through the air18. In addition, the wideband antenna shall suffer higher if the mutual coupling between elements is high. It is typically suggested that the inter-element distance should be at least half-wavelength to counter issues that emerge through mutual coupling. The arrangement of antennas for MIMO shall have to deal with the tradeoff between mutual coupling and antenna radiation characteristics. A good MIMO characteristic shall need at least mutual coupling isolation of 20 dB. There are several decoupling mechanisms. A significant quantum of research is present in the literature for improving the antenna isolation characteristics22. The presented antenna does not employ any additional decoupling mechanism for improving the isolation characteristics. State-of-the-art research in MIMO antenna design has been recently reported24. The literature highlighting bandwidth and gain enhancement techniques for MIMO antenna can be found in30.The multiple-input multiple-output (MIMO) technology enables enhanced channel capacity and moderates multipath fadingA novel antenna design structure which was designed with the different iteration of the shape to achieve the desired frequency band of operation.Proposed MIMO antenna designed by engraving the single side of the PCB where the ground plane does not require the engraving process. This will reduce the overall fabrication cycle for the antenna.The values of the cross-port isolation are better than 15 dB which provides good isolation between antenna elements. Hence, due to the excellent inter-element isolation, the antenna performs very well as a radiator regarding MIMO diversity performance.This structure offers the dual band of the operation with maximum of 4.2 dBi of the gain 2.8 GHz of the bandwidth.The antenna structure and the different approach can help to design the frequency selective dual band antenna. Our antenna design approach helps to identify the resonating frequency bands between 5.2 to 5.6 GHz where the first frequency band are almost same in all the iteration.The patch antenna provides ease in integration and manufacturing. Mobile communication antennas significantly suffer from space-constrained environments. The surface mountable antennas provide viability of the placement in such conditions. The patch antennas usually suffer from less bandwidth and moderate gain issues because of the dielectric material loss at high frequencies above 1 GHz. The presented study focused on high inter-element isolation, good antenna gain and improved MIMO characteristics. A quad-element MIMO antenna is aligned orthogonally to contribute to targeted applications to achieve the desired antenna parameters. The radiating elements of planar resonators are kept in anti-parallel mode to improve diversity parameters significantly. The antenna presents its usage in NR FR-1 5G communication and WLAN applications. The novelties of the structure achieved in this manuscript are as follows:3 or the electrical dimensions of 1.06 λ × 1.06 λ × 0018 λ at a lower frequency. The quality of a patch antenna can be increased in several different ways: by using a thick substrate, cutting a resonant slot inside the patch, using a low dielectric substrate, configuring multiple resonators in a stack, using different impedance matching and feeding methods, and by employing slot antenna geometry. Improving one attribute often worsens the other; this is especially true when it comes to antenna bandwidth and size, and therefore the careful approach to design evolution with necessary computational checks at every step is required in antenna design.The proposed quad-element MIMO antenna is designed on AD255C Rogers laminate. The AD255 has low-lossy characteristics with a loss tangent of 0.0014 and a dielectric constant 2.55. It has a very high copper peel strength of more than 10  pli, a good coefficient of thermal expansion in order 34 ppm/°C, a strong tensile strength of 55.8 MPa and radiating element copper profile of 35 oz. The geometric design of the proposed single-unit antenna is exhibited in Fig. 11 results improved, but the resonance peak did not occur in the desired wireless frequency. After considering and reviewing a mountain of literature on sliding the resonant peak by carving out the patches in microstrip patch antenna, and thus more carving out occurring like a driven element in the yagi-uda antenna, the width of the deformed rectangular patch was kept wider on the right and narrower on the left. This concept successfully produced the desired s parameters, and its resonating point was in the NR FR1 and WLAN bands. To fine-tune and optimize the design, dividing the square area into wider elements was determined to achieve optimized results in the preferable frequency band. For all four phases of the design, the bottom view of the antenna states that the full ground plane of the metal has been placed with no further modifications. Figure 11 parameters.The proposed antennas have undergone four phases of development, and the fourth and final phase has been demonstrated to produce the desired results. Phase 1 consists solely of the carved-out rectangular portion for improved bandwidth and s-parameters. In the second phase, square sections were added to the bottom right corner of the carved-out rectangular patch, as optimal reflection results were not quite attained. The phase 2 S11 parameter rests in its capacity to measure the quantity of power reflected by the transmitter from the antenna. This parameter denotes the degree to which the antenna and transmission line or source impedance are matched. A low S11 value indicates a good impedance match, which means that most of the power is transmitted from the transmitter to the antenna, and very little is reflected. A high S11 value, on the other hand, signifies an impedance mismatch that results in a significant quantity of power being reflected in the transmitter. MIMO systems require a strong impedance match for efficient power transfer and signal integrity. When the S11 parameter is small, it ensures that the majority of the transmitted power is effectively radiated by the antenna instead of being squandered as reflected power. This results in increased signal strength, decreased interference, and enhanced overall system performance.In antennas, the significance of the S11 parameter aids in determining the antenna's operating bandwidth. Analyzing the S11 characteristics across a range of frequencies makes it feasible to identify the frequency bands in which the return loss is acceptable. This information facilitates the selection of appropriate frequency bands for MIMO operation and the design of filters or other components to enhance antenna performance within those frequency bands. The S11 parameter contributes to the fine-tuning of patch antenna design. Adjustments to the antenna size, substrate type, feeding structure, and other factors may be made using the S11 values to optimize performance and obtain desired features such as resonance frequency, radiation pattern, and gain, and Fig. In addition, the S11 and frequency for all four phases is illustrated in Fig. 11 is nearly 0 dB across the entire band, with one good resonance of approximately − 30 dB outside any of the wireless frequencies we are concentrating on. Phase 2 yields improved reflection coefficient results, but not in the desired wireless frequencies. Figure 11 resonance are occurring in the desired wireless frequencies. Though a − 20 dB resonance is less desirable than a − 30 dB resonance, some of the applications in these wireless bands where moderate levels of reflections are tolerable for IoT devices, non-critical wireless communication systems, home Wi-Fi networks, low-power wireless sensor networks, and indoor communication networks so long as the system provides adequate communication quality.The relationship between SA parametric study was conducted to enhance an antenna's design and optimization processes while addressing the issue of resonant frequency and improving its performance in terms of return loss. The study examined various parameters, including slot lengths and widths, to determine their impact on the antenna's return loss and impedance bandwidth. To better understand the effects of changing parameters on the antenna’s performance, only one parameter was altered at a time while keeping the others constant. Several parameters influence the resonant frequencies and matching of the antenna and Fig. All the parameters in a patch antenna control the properties of the antenna, such as length, width, height, and permittivity. Here, a parametric study of the length and width of different slots is given as properties such as resonant frequency, input impedance, return loss, and radiation patterns were the focus of this MIMO antenna structure. The slots length and not the width heavily influence the resonant frequency, while the relationship between the length of the slot and resonant frequency can be driven as per Eq. :1\documeThe width of the slots affects the input impedance and radiation patterns of the antenna. It has been theorized that the wider the patch, the lower the input impedance. The slot's width predominantly determines the coupling between the slot antenna and electromagnetic radiation. A narrower slot width enhances the intensity of the electric field within the slot, resulting in a greater input impedance. It is because a narrower slot restricts the passage of currents along the slot margins, resulting in a greater concentration of electric field and a greater impedance.On the other hand, the slot length influences the antenna's resonant frequency. The length of the slot, which functions as a resonator, determines the wavelength at which the antenna emits or receives electromagnetic radiation efficiently. The antenna can be tuned to resonate at various frequencies by altering the slot length. In contrast to the slot width, the effect of slot length on input impedance is relatively minor. The relation between width and impedance for the rectangular-like patch can be approximated by given Eq. .2\documeInput impedance is directly associated with the return loss. The width parameter is heavily effective in improving input impedance and, thereby, return loss because when the slot width of an antenna is increased, the effective aperture of the antenna also increases. This results in the antenna capturing more electromagnetic energy, which leads to higher power reception. Consequently, the reflected power decreases, and the return loss improves. It means that increasing the slot width usually leads to a reduction in the magnitude of the reflected power and an improvement in the return loss.On the other hand, decreasing the slot width of an antenna results in a decrease in the effective aperture of the antenna, leading to lower power reception. It can cause more energy to be reflected, increasing the magnitude of the reflected power and worsening the return loss. Figure Furthermore, modifications made to the slot width or length can have a cascading effect on other antenna parameters, such as bandwidth, radiation pattern, and efficiency. Thus, it is essential to consider all pertinent parameters and their interrelationships when designing a slot antenna to ensure optimal performance. The proposed resonator design employs the four MIMO elements placed orthogonally at the corner of the top layer, as illustrated in Fig. A common configuration in MIMO systems involves placing four antenna elements at the four corners in an orthogonal position. Placing the antenna elements at the four corners in an orthogonal position is advantageous, enabling spatial separation and maximizing the spatial diversity gain. Positioning the antennas at different corners allows the signals to undergo varying propagation paths and fading conditions, ultimately improving the overall system performance. The orthogonal placement of the antenna elements involves arranging them so that their polarization or radiation patterns are orthogonal to each other. This arrangement allows for more independent spatial channels and reduces interference between the antenna elements.Consequently, it enables higher data rates and improves the communication reliability of the system. The proposed resonator design employs the four MIMO elements placed orthogonally at the corner of the top layer, as illustrated in Fig. Figure It is found that the measured results are also matched with the simulated structure and show the two similar operating bands of the resonance. There is a minute difference between simulated and measured results. There multiple reasons between simulated and measured values (a) The instrument errors, (b) fabrication tolerances and (c) ideal conditions set by software simulator while numerically computing the parameters. A few of iterations were carried out to confirm the presented results. The surface currents at targeted frequencies are shown in Fig. Figures 31. This method makes use of the S-parameters for the calculation of the ECC.In Eq. , ECC is Calculations for ECC are shown in Fig. 31 is feasible to confirm that the MEG is present. As mentioned in31, the total number of ports in the current architecture is M. The η denotes the radiation efficiency of the present MIMO structure rad. If you want the optimum performance from the device’s diversity feature across all ports, you must ensure that MEG is set to − 3 dB. In addition to this, the level differences between the two ports must be equal to 0 dB, as shown in Fig. If you put two isotropic antennas at a location where there is no background noise, the total average power that those antennas will receive will either be the same as or less than the average power that a diversity antenna would receive if it is placed in a location where there is background noise (or interference). For instance, the increased performance of a MIMO antenna could be affected by external variables such as the environment in which it is placed. Applying Eq.  as recom33. Figure The Total Active Reflection Coefficient, which is more often referred to as simply TARC, is capable of giving an accurate evaluation of radiation performance as well as frequency response when it is applied to a large number of ports. To compute it, take the square root of the total reflected power and divide that amount by the total power that was incident on the item. This will give you the value you're looking for. The solution will be found in the result. The Total Active Reflection Coefficient, sometimes known as TARC, is a statistic that assesses how well a MIMO system bends light. This strategy considers both the random signal pairings that might take place and the mutual coupling that can take place across different networks. It is feasible to analyse the classification of reflected and incident waves using Eq.  in the p32. As shown in Fig. In assessing the MIMO performance of the chosen THz antenna, the Channel Capacity Loss is an additional significant component that must be considered (CCL). The amount of data that can be sent across a channel at a certain speed without experiencing a substantial loss is the channel’s capacity loss. To successfully signal that information has been sent, the rate must be less than 0.5 bits/second/hertz while using a MIMO system that has been created efficiently. Calculating the CCL parameter may be done using Eqs. –10, as sThe antenna diversity gain (DG) provides the improvement in signal-to-noise ratio (SNR) when the MIMO antenna diversity scheme is established. In the existence of the MIMO system, the SNR is enhanced and subsequently the signal reception. This improves the reliability of the communication system. The DG can be computed from calculated values of ECC as shown in Eq.  the valuThe antenna is compared with other antennas in the literature to show its utilization in targeted frequencies as exhibited in Table We show a four-port MIMO antenna that has a high level of isolation. The antenna's primary purpose is to cover the n48 band of Frequency Range-1 (FR-1), which operates in TDD duplex mode. After achieving the size reduction of a single antenna unit by optimizing the placement of slots and extended arms, the quad-antennas are arranged orthogonally to create antenna diversity. The antenna has resonant frequencies of 3.56 GHz and 5.28 GHz, with a 2:1 VSWR fractional bandwidth of 1.82% and 2.12%, respectively. The effectiveness of the suggested resonator is 88.34% for lower bands and 79.28% for higher bands, accordingly. Due to the excellent inter-element isolation, the antenna performs very well as a radiator regarding MIMO diversity performance. The figures for the envelope correlation coefficient are − 0.005, and the channel capacity loss is 0.1 bits/sec/Hz. The overall active reflection coefficient is − 24.26. High directivity and cross-pol isolation are both helped by the ground plane's complete profile. The antenna has a gain of 4.2 dBi and, accordingly, 2.8 dBi, which is sufficient to meet the requirements of the intended applications. Reported results in the proposed antenna strongly apply to the device used in 5G and WLAN Communication.


In this paper, a 12-port MIMO antenna system for 5G/WLAN applications is proposed. The proposed antenna system consists of two types of antenna modules: an L-shaped antenna module covering the C-band (3.4–3.6 GHz) for 5G mobile applications and a folded monopole module for the 5G/WLAN mobile application band (4.5–5.9 GHz). Each two antennas form a pair, six pairs in total, forming a 12 × 12 MIMO antenna array, and the elements between the antenna pairs can achieve an isolation of 11 dB or more without additional decoupling structures. Experimental results show that the antenna can cover the 3.3–3.6 GHz and 4.5–5.9 GHz bands with an overall efficiency greater than 75% and an envelope correlation coefficient less than 0.04. Finally, the one-hand holding mode and two-hand holding mode are discussed to demonstrate their stability in practical applications, and the results show that they still exhibit good radiation and MIMO performance when operating in both modes. The fifth generation of mobile communication (5G) has been officially put into commercial use, and one of its main features is the high-capacity and high-rate transmission of information . MultiplCurrently, different countries and operators use different frequency bands for 5G communication. For example, LTE band 42 (3.4–3.6 GHz) and LTE band 43 (3.6–3.8 GHz) have been certified and adopted by the European Union [However, for MIMO array antennas applied to cell phones, mutual coupling between MIMO units will be unavoidable due to the limited space of cell phones. Severe mutual coupling will lead to the degradation of isolation and seriously affect the channel capacity of MIMO systems . Since tTo meet the needs of mobile communications, there is a large body of literature on designing MIMO antennas for 5G smartphone applications ,25,26,27Based on the above status, this paper proposes a 12-port highly isolated broadband MIMO antenna system for 5G/WLAN applications. In order to support multiple communication modes and improve channel capacity, the proposed antenna system is made up of two different types of antenna modules. One is an L-shaped antenna, consisting of an L-shaped metal patch and a T-shaped slot, which covers the C-band (3.4–3.6 GHz) for 5G mobile applications. The second module is a folded monopole consisting of a meandering metal patch and a rectangular slot, and each antenna unit of this module covers 4.5–5.9 GHz used for 5G and WLAN mobile applications. Simulations (obtained by HFSS) and observations corroborate the performance of the proposed 12-port MIMO system.3, which is suitable for 5.5-inch smartphones. The metal ground (140 × 70 mm2) is printed on the back side of the FR4 substrate. The precise construction of the antenna components (Ant1 and Ant2 as examples) is depicted in The proposed 12-port antenna structure is shown in 2, as shown in Each antenna element in the L-shaped antenna module is supplied via a 50-ohm L-shaped microstrip feed line linked to the ground plane via an SMA connection. On the bottom of the substrate, a T-shaped slot radiator is etched where the printed metal floor corresponds to the L-shaped antenna. This antenna, unlike traditional closed-slit antennas, employs an etched T-shaped slit radiator with open branches. T-slits boost antenna capacitance by inserting I-shaped opening branches into the center of the U-slit. The overall size of each element of the L-shaped antenna is 3 × 8 mm2.In the folded monopole module, each antenna element consists of a rectangular ground-plane slot and a meandering metal strip. The folded monopole is printed on the FR4 substrate’s top layer. Each folded monopole has six 50 vertical feed strips and six horizontal feed strips for tuning. The antenna element may be set to the required frequency range by altering the width and length of the feed strips. The entire size of the folded monopole element encompassing the high-frequency range for 5G mobile applications and the WLAN band (4.5–5.9 GHz), as shown in In order to study the working mechanism of the proposed antenna model, a parametric study of the individual antenna elements of the two modules was carried out using the electromagnetic simulator HFSS. Since the six antenna elements in each module have the same geometry, the parameters of Ant1 and Ant2 are used to analyze the excitation of the proposed antenna system in the low- and high-frequency bands. To verify the feasibility of the proposed antenna system, we fabricated and tested the proposed antenna model. The proposed array antenna design was investigated in depth by the simulation software HFSS version 20. To understand how the two-antenna module tuning mechanisms affect the impedance matching and operating bandwidth of individual antenna array elements, a parametric study was conducted with Ant1 and Ant2 as examples. It is worth noting that when one of the parameters is adjusted, the other parameters will remain unchanged.2 at the front of the horizontal microstrip line. In the L-shaped antenna module, unlike the conventional L-shaped metal patch, the proposed patch structure in this paper adds a small metal patch of 0.5 × 0.5 mm7 and L12. 7 on the reflection coefficient by expanding the slot width from 2 mm to 8 mm, resulting in a resonant mode at 5.0 GHz. The slot width was lowered for higher frequency bands by reducing the width of W7, which narrowed the slot width and shortened the current flow. In summary, W7 may be used to adjust the antenna element’s center frequency in the appropriate frequency band. 12, another parameter that influences the resonance frequency, is altered from 3 mm to 9 mm. The resonant frequency is tuned to a higher band and the operating frequency is shifted to a higher frequency. Based on the above results, the W7 width is set to 6 mm and the L12 length is set to 12 mm, and the frequency band can be adjusted to the desired value.A parametric study was performed in the folded monopole module to investigate and analyze the effect of different lengths and widths, including WECC is one of the main indexes to evaluate the performance of the MIMO antenna array. The concept of ECC is mainly used to quantitatively describe the mutual independence of the transmitted signals between different antenna units in the MIMO antenna array: the smaller the value of ECC, the greater the mutual independence of the transmitted signals between different antenna units in the MIMO antenna array. In MIMO antenna design, the value of ECC is required to be less than 0.5 . The mosAs can be seen in In this section, a preliminary analysis of the proposed twelve-cell antenna’s peak gain and overall efficiency findings is performed. The radiation efficiencies of the L-shaped antenna module and the folded monopole module (Ant1 and Ant2 as examples) are depicted in In this subsection, the directional maps (E-plane and H-plane) of the proposed 12-port antenna array with L-shaped and folded monopole elements measured at two representative frequencies (3.5 GHz for the low-frequency band and 5.5 GHz for the high-frequency band) are discussed and analyzed separately, considering the similarity of the measured radiation, with the results of Ant1 and Ant2 as representative ones. In this section, the effect of the user’s hand on the performance of the proposed antenna is investigated. There are two common usage modes of embedded smartphones, namely, talk mode (SHM) and data mode (DHM). The hand configurations of the proposed antenna design in two different modes of SHM and DHM are shown in In the talk mode (SHM), the reflection coefficients of the L-shaped antenna and the folded monopole module are not significantly affected by the user’s hand, as shown in The functioning of the data mode (DHM) is depicted in the figure; as shown in To highlight the advantages of the proposed antenna array, the comparison results of the same type of antennas are given in In this paper, a 12-port highly isolated broadband MIMO antenna system for 5G/WLAN applications is presented. The proposed MIMO antenna consists of an L-shaped antenna module covering the C-band (3.4–3.6 GHz) for 5G mobile applications and a folded monopole module covering the high band for 5G mobile applications as well as the WLAN band (4.5–5.9 GHz), where each antenna module consists of six identical antenna elements. In addition, the antenna pair composed of two antennas can achieve isolation higher than 11dB without adding additional decoupling facilities, obtains wideband characteristics, high isolation and low ECC, and also has good diversity performance, and therefore can be used in modern mobile terminals.


By stacking the DRA with a one × three array of H-SRR unit cells, a 30 dB reduction in the mutual coupling level is attained without compromising on the antenna performance. The corresponding mutual impedance of the MIMO DRA is better than 30 dB over 5.9–6.1 GHz operating bandwidth. The proposed design has a DG of 10 db, ECC < 0.02, CCL < 0.02 bits/s/Hz, and an MEG of 0 dB. The overall design has a promising performance, which shows its suitability for the target wireless application.A single negative metamaterial structure with hexagonal split-ring resonators (H-SRRs) is inserted within a two-port multiple-input multiple-output (MIMO) dielectric resonator antenna (DRA) in order to achieve a reduction of mutual coupling between closed multiple antenna elements. Between closed, tightly coupled, high-profile antenna elements, the single negative magnetic inclusions (H-SRRs) are embedded. By incorporating magnetic structures within antenna elements, the mutual coupling is significantly diminished. Mutual coupling reduction is attained by inserting an array of hexagonal split-ring resonators between the inter-spacing elements. An operative approach for the reduction of the mutual coupling between two  During the last decade, with the development of communication systems, the demand of the capacity and data rates for internet and other services  has highly increased. This technological development has considerably increased the number of subscribers. To fulfill the new requirements, the concept of multiple antennas (MIMO) was proposed. The main challenge of the multiple antennas device/system is the size constraint. Electronics system design engineers are therefore focusing on the miniaturized transmitting/receiving (Tx/Rx) devices, while maintaining a high data rate, enhanced channel capacity, and better coverage of the signal . There iIn ,13,14,15aterials . For thiaterials ,9, high aterials , complematerials ,12, and aterials . The desaterials . Such staterials ,19,20,21aterials . This isaterials . In thisIn , a spira designs .Mutual coupling highly depends on the spacing between the antenna elements. Closely spaced antenna elements offer high coupling with reduced port isolation. This causes unwanted effects on antenna features like reduced ECC and low DG, TARC, and MEG. Contrary to closely spaced antenna elements, increasing the inter elements distance reduces the mutual coupling, but it causes an enlarged size and affects the radiation pattern. Within a compact device, reduction in mutual coupling in MIMO antennas is a major challenge. As the proposed design is working at a lower frequency, handling the mutual coupling at this range of frequencies is even more challenging. We need techniques which are helpful in reducing the mutual coupling while maintaining a compact size and high performance. Thorough investigation of multiple techniques reveals that the deployment of metamaterials (CSSRs and SSRs) among the antenna elements is the ultimate solution to this issue. These uniquely structured materials have the potential to highly reduce the mutual coupling while maintaining a good overall performance .The main objective of this work is to combine H-SRRs for first time with RDRA MIMO antenna elements for mutual coupling reduction. H-SSRs offer wideband response with a compact size. Previously, this technique was combined with cylindrical DRAs. It is well known that RDRA has different operating modes and shape than the cylindrical DRA. Thus, integrating H-SRR with RDRA is a real challenge. Careful design and optimization have enabled a successful deployment which results in a significant reduction in mutual coupling (from 6 dB to 30 dB).The rest of the paper is organized as follows: 1 is the center-to-center distance between the DRs. Each DR of the proposed design are excited by a coaxial probe feed, which is placed at the bottom of the ground plane. For the isolation enhancement, a metamaterial structure with three pairs of hexagonal-shaped split-ring resonator (H-SRR) is introduced between the DRs. The detailed dimensions of the proposed design are listed in The evaluation process of the MIMO DRA is depicted in In (2), In this case, the DR resonates at 5.2 GHz with good impedance bandwidth and matching. The simulated S11 are depicted in The unique characteristics of a metamaterial shield are employed in the proposed MIMO DRA presented in this section. The metamaterial structure comprises a unique unit cell (SRR) having dimensions “r (3.9 mm)”, “g (0.5 mm)”, and “c (0.6 mm)”, where r is the radius, g is the split width, and c is the SRR width, as shown in To achieve the optimum choice of H-SRR with improved results for the proposed design, a parametric analysis is carried out on important parameters of the SRR. Different parameters of the SRR have different effects on the performance of the proposed design. These key parameters include the width of the splits and the width of the metal of the inner sides and outer sides.As shown in Increasing the gap split decreases the capacitance, which in response increases the resonance frequency. Based on the return loss response, the width of the split is attained at 0.5 mm.The width of the SRR also plays an important role in attaining the desired resonance frequency. As shown in In this section, we present the final simulated and measured results of the proposed design, providing a comprehensive evaluation of its performance. The key parameters analyzed include reflection coefficients, mutual impedances, gain, and radiation patterns across all operating bands. The mutual impedances between the antenna ports are also examined to evaluate the level of interference and coupling between the elements.The proposed MIMO-DRA, integrated with a one × three array of an H-SRR unit cell, is fabricated and its performance is measured. The prototype photograph of the proposed MIMO-DRA with H-SRR is depicted in The gain characteristic is a vital indicator of an antenna’s performance and its ability to radiate and transmit power effectively. The gain of the proposed design across the operating bands is evaluated through simulation and measurement, as depicted in Surface current distribution is helpful in explaining the overall behavior of the proposed design. To assess the MIMO and diversity performance of the proposed antenna, key parameters such as the envelope correlation coefficient (ECC) and diversity gain are evaluated.The ECC is a crucial metric in MIMO antenna systems as it quantifies the correlation or isolation between the different branches of communication. In this study, the ECC is computed using the radiation pattern of the proposed MIMO antenna and can be expressed using Equation (7). The measured pattern is noticed at three frequencies . Then, by using Equation (7), the measured ECC at three points is noticed. The final response is drawn by curve fitting. It is generally desirable to have an ECC value below 0.5 for satisfactory MIMO performance ,23. FiguThe diversity gain (DG) is another important parameter that evaluates the enhancement provided by a MIMO system compared to a single antenna system. The DG is calculated using Equation (8), which quantifies the improvement in signal quality achieved through diversity techniques. Channel capacity loss (CCL) is another important MIMO parameter. CCL is important in showcasing the effectiveness of the proposed design throughput. The smaller the value of the CCL, the better the data transmission. A 0.4 bits/s/Hz CCL value is considered a good one for good data transmission. Equations (9)–(12) explain the basic parameters which help to determine the final CCL response of the proposed design .(9)CLosMean effective gain (MEG) is another important MIMO antenna design parameter. It is the ratio between the diversity antennas’ received power and the isotropic antenna’s received power . It showThe total active reflection coefficient is also an important MIMO antenna parameter. It is the ratio between the square root of the total power reflected and the total power incident. It is helpful in the determination of the effective operating bandwidth of a MIMO antenna system. The TARC of a MIMO antenna system is determined with the help of the following equations :(15)TARCFor good communication, the TARC of a MIMO system should under 0 dB . Thus, aA comprehensive comparison between the proposed work and recently published studies is presented in Mutual coupling between closely placed dielectric resonator antennas on a multiple antenna system was investigated in the present work. A single-negative magnetic (MNG) H-SRR band gap decoupling structure was examined. Detailed simulation and experimental measurements were performed as well. The magnetic insertion has shown the effectiveness in terms of isolation enhancement and suppressing the displacement current. From the simulation results, a 6 dB isolation was achieved without H-SRR insertion and 27 dB additional isolation was achieved at the resonance frequency by employing the H-SRR. The isolation enhancement has resulted in a recovery of the multiple antennas pattern, which is clear from the pattern measurement performance. The computed results confirm that implementing the H-SRRs between the antennas offers an excellent coupling reduction. The benefit of the proposed design is its small electrical size (λ/12) and simplicity of fabrication as compared to other meta-structures. The overall design compactness and novel decoupling structure emphasizes the significance of the proposed design for real-time wireless applications.


A unique high gain antenna array with a 3D-printed dielectric polarizer is proposed. The packaging of the antenna array feeding structure is eliminated by aggregating the feeding network in between the antenna elements. This has a significant advantage in maintaining neat and symmetric radiation characteristics with low cross-polarization levels. The proposed structure combines two elements in one feeding point to reduce the array distribution feeding points of a 4 × 4 antenna array from 16 to 8 points. The proposed antenna array structure is extremely low in cost and can be used as either a linearly or circularly polarized one. The antenna array achieves a gain of 20 dBi/dBiC in both scenarios. The matching bandwidth is 4.1%, and the 3-dB Axial Ratio (AR) bandwidth is 6%. The antenna array uses a single substrate layer without the need for any vias. The proposed antenna array suits well various applications at 24 GHz, while maintaining high performance metrics, and low cost. The antenna array can be easily integrated with transceivers due to the use of printed microstrip line technology. While we can have higher speeds at mm-Wave frequencies, the realization of the physical layer becomes more challenging7. The main disadvantage of operating at higher frequencies is having a higher path loss for the wireless propagating electromagnetic waves once compared with lower RF frequencies. To compensate the path loss, one might suggest increasing the power amplifier gains in the radios. The main issue with this solution is not only that it will drain more power from the supply and cause more heating, but it will also make the devices bulky by accommodating the heat sinks and the required cooling apparatus. For mobile devices, it would be impractical as it will drain the device battery very quickly. A suggested remedy is to use high directivity antennas that will focus the energy towards the communicating entity, this will compensate for the path loss effect and relax the design requirements of the power amplifiers15.In wireless communications, the channel capacity is proportional to the available bandwidth according to Shannon’s limit. The more bandwidth we have, the higher the capacity of the wireless channel is. As such, a higher data rate can be achieved. By migrating the operation to higher frequencies, such as the anticipated mm-Wave frequencies. The absolute available bandwidth would be significantly larger than the typical RF frequencies. Therefore, higher speeds of wireless communication become attainable20. The use of short-range radars for health care applications 21, automotive radar sensors, and motion detectors has become ubiquitous26. Moreover, the developments in wireless connectivity led to the invention of various internet of things technologies. The Internet of Things encompasses plenty of applications. An antenna is an integral part of any IoT communicating device. The performance of these antennas is a crucial factor for the whole system performance. Various antenna structures have been proposed in the Industrial Electronics33, IoT and Sensors literature41. In Ref.42 a patch antenna with a fence-strip resonator was realized for smart homes IoT communication. For such communication the antenna radiation pattern have to be Omni-directional. In Ref.43 a unique glasses frame antenna for IoT communication was realized. A programmable beam scanning antenna without phase shifters was proposed for IoT relay communication in Ref.24. In Ref.26 a shark-fin antenna was realized, the antenna is to be used for future railway communication systems. In Ref.23 a multiband printed smartwatch antenna was proposed, the antenna enhanced the number of frequency bands and improved the Omni-directivity. In Ref.22 a microstrip patch antenna was employed in a structural health monitoring (SHM) system to measure structural strain. In Ref.44 an electronically steerable parasitic array radiator was used in a dense wireless sensor network. In addition to that, the modelling of antennas is essential in the design process, as an example but not limited to, since the existing antenna equivalent models are inflexible because they assume rectangular antenna contour, a hybrid-equivalent surface-edge current model was proposed in Ref.45 to overcome the limitation of the existing models, these models are very useful for vehicle to everything (V2X) communication. Metasurfaces and dispersion engineering techniques can be found to be very useful for various applications as well, Metasurfaces can be utilized to manipulate the characteristics of the propagating waves effectively47.Operating at mm-Wave frequencies is useful for radar and sensing applications as well. The higher the radar frequency of operations is, the higher the resolution that can be achieved. Several works proposed the use of the 24 GHz mm-Wave band for radar applications49. In this article we present an extremely low-cost antenna array that can be used as either linearly or circularly polarized one. The antenna array has a realized gain of 20 dBi in both scenarios. The matching bandwidth is 4.1% and the AR bandwidth is 6%. The antenna only uses a single substrate layer with no need for any vias. To switch to circular polarization operation, a 3D printed dielectric polarizer is used. The proposed antenna array suits well various applications at 24 GHz, while maintaining high-performance metrics, and extremely low cost. The antenna can be easily integrated with transceivers due to the use of printed microstrip line technology.The polarization of the antenna is usually determined by the application, for example in satellite communication systems due to the difficulty of alignment, a circular polarization is necessary to avoid any polarization mismatch loss. Radars uses a linear polarization frequently, but they can still use a circular polarization as well50. Reflectors, and lenses are well known for their ability to provide high gain51. However they are very large due to their focal length requirement, which makes them harder to integrate. Moreover, they are not considered low-profile solutions. Printed technology is well suited for highly integrated systems, and they are well known for being low profile. Planar arrays can be used to increase the gain performance. By increasing the number of elements in an antenna array, the gain increases proportionally, the rule of the thumb is that by doubling the number of elements the gain increases by 3 dB. However, the issue is that by increasing the number of elements, the associated feeding network size would increase in a proportional manner. The larger the feeding network is, the lager the incorporated loss in it. The total loss in a printed feeding structure is the sum of the dielectric, conduction, and radiation losses. Radiation loss can be reduced by minimizing the radiation produced by the feeding network, ideally a closed structure would suffice to eliminate the radiation loss completely. The dielectric loss can be reduced by using a material with a low loss tangent, ideally vacuum (air). Therefore, closed metallic waveguides are considered to be very efficient feeding structures as they eliminate both radiation and dielectric losses, however they are very expensive and harder to integrate with printed circuits due to their feeding waveguide transition requirement53. Moreover, by shifting to higher frequencies, the dimensions become smaller and harder for the milling machines to realize. Accordingly, this makes them more prone to tolerance errors, and more challenging to be realized by milling machines. Here, we show that we can maintain a decent efficiency and high gain by using printed technology, while at the same time keeping the structure unpackaged, via-less, and only using a single substrate layer. Figure 50 we showed that such a printed open microstrip line stub can be used as a radiating element where the radiation from the fringing fields can be utilized. As in Fig. 54.At higher frequencies, the gain of the antenna is essential to compensate for the path loss; there are several ways and various structures that can be used to achieve high gain performance metric50. To tackle this issue, the feeding network design procedure in Fig. L as long all the lines lengths are multiple of an odd integer of the guided quarter wavelength. This procedure is very useful where it allows the use of narrow microstrip lines that can fit easily between the radiating elements. Moreover, it minimizes the whole parasitic radiation from the feeding structure due to the use of thin lines which are aggregated between the elements. Further detailed analysis of this concept and comparison cases with packaged structures can be found in Refs.56. Figure 57. Packaging is a good choice; however, it is not easy to fabricate, and it is more expensive. In the suggested design, the packaging of the antenna array feeding structure is eliminated by aggregating the feeding network in between the antenna elements, this has a significant advantage in maintaining neat and symmetric radiation characteristics with a decent gain in the range of 20 dBi, further details about this technique can be found in Ref.50. Figure 58).The typical procedure to design a feeding network for a 2D array is by using power dividers and quarter wavelength transformers to match the parallel combination of the antenna elements to the desired system impedance. This procedure usually results in transmission lines with different characteristic impedances. Due to the limited space between the antenna array elements, the wide microstrip lines cannot be accommodated, especially in the aggregated configuration. Moreover, very wide microstrip lines can have significant radiation loss65. For example, they can transform a linearly polarized wave to a circularly polarized one and vice versa. Figure 67. The operation of the dielectric polarizer can be explained by decomposing an incident plane wave with 45° angle into two components, parallel and perpendicular or (TE and TM components) as shown in Fig. 68, consequently, the phase difference and the axial ratio can be calculated as in (7–8).Simply put, electromagnetic polarizers are structures that are used to transform the wave polarizationo × 3.6 o and the polarizer height is 1.87 o. Figure Table xz and yz planes. The radiation pattern shows an LHCP wave in both planes with side lobe level way below − 14 dB. Figure 11 in both cases are almost equal indicating a transparent polarizer.Figure The 20 dBi gain, and the 90% efficiency of the antenna array in both scenarios makes the antenna very attractive for 24 GHz applications . The antenna can be used effectively in either mode (CP or LP) depending on the application requirements. While the radiation characteristics encompasses a neat directive pencil beam, the side lobe level is well reduced below − 14 dB in both scenarios. The solution is highly suited for integration with PCB technology. The structure is extremely low in cost where it only uses a single substrate layer with no vias.θ and Eϕ) for the linear polarization, and  for the circular polarization. Figure LP = Eθ − Eϕ), and the CP case is given by (ΔCP = ELH − ERH). As expected the circular polarization gain is independent of the azimuthal angle, which makes it immune to misalignment polarization miss match. Also the cross polarization level in the CP case fluctuates from 16.4 to 22.6 dB with the azimuthal angle (i.e. within a 6.2 dB margin). This feature of CP antennas makes it very desirable for several applications which needs an aligned line of sight wireless communication link. On the other hand, the LP antenna shows that the electric field gain depends highly on the azimuthal angle. . In the E-plane, the co-polar component is Eθ, and the cross-polar component is Eϕ, and vice versa in the H-plane. Therefore, with the azimuthal misalignment the gain drops from one component and increases in the other component, and the cross polar level can vary from 29.6 to 0 to − 25 dB, the change in the sign represent the switch from one principle plane to the other. Despite the fact that LP antennas are prone to polarization mismatch loss, they well fit several radar applications, this is due to the fact that reflected signals from objects maintain the same polarization (i.e. Linear), and allows the use of one linear antenna for both the transmitter and receiver circuitry which are usually connected through a circulator. On the other hand, in the CP case, the reflected signals from objects flip their polarization . As such, the transmitting antenna cannot detect the reflected signal, and this will require another antenna with the opposite polarization to detect the reflected signal.It should be clear that there is a slight azimuthal misalignment of the antenna due to the visual alignment of the antenna in the chamber while performing the measurement. The misalignment is noticed from the measured cross-polarization component, which has some contribution from the co-polar in that case. That is why a peak is shown on the broadside of the cross-polarization. Figure 69, an 8 × 8 patch antenna array can only provide 18 dBi of gain.A comparison with other works is shown in Table The proposed structure only uses a single substrate which makes it extremely low in cost. It doesn’t require any vias, which simplifies the fabrication process significantly. Such simplification in the fabrication is very beneficial, especially at millimeter wave frequencies where the structure/vias dimensions become very small and very susceptible to tolerance errors. Furthermore, the proposed structure uses a single substrate with a MSL feed, which makes it very easy to be integrated with transceiver circuitries. Other works as indicated in Table A low-cost antenna array operating at 24 GHz was demonstrated, the packaging of the antenna array feeding structure has been eliminated by aggregating the feeding network in between the antenna elements. The proposed antenna array had an extremely low cost and could be used as either a linearly or a circularly polarized antenna array. The antenna array had a realized gain of 20 dBi in both scenarios. The achieved matching bandwidth is 4.1% and the 3-dB axial ratio bandwidth is 6%. The antenna array only used a single substrate layer with no need for any vias. To switch to circular polarization operation, a 3D printed dielectric polarizer was used. The proposed antenna array demonstrated well suitability for various applications at 24 GHz, while maintaining high performance metrics, and extremely low cost. The antenna array could be easily integrated with transceivers due to the use of printed microstrip line technology.


The antenna structure incorporates three U-shaped slots in the ground plane, forming a defected ground structure. The microstrip line is exclusively present on the front side plane. The antenna is printed on a substrate made of a ceramic-filled PTFE composite with a size of 20 mm × 21 mm × 0.76 mm and a dielectric constant of 3. The proposed antenna is analyzed using the characteristic mode analysis based on the method of moment and simulated by an electromagnetic simulator based on the finite element method. A multiband antenna is fabricated and tested to validate the proposed antenna performance. The simulation and measurement results reveal that the antenna exhibits good input impedance bandwidths of S Multiband printed antennas have been the research priority with this development. Printed antennas have the advantages of low profile, low cost, lightweight features, easy manufacture, and integration with microwave monolithic integrated circuits (MMICs)4. The usage of DMS and DGS for realizing the design of a mirror stairs microstrip multiband antenna that covers the range of 2–17 GHz is explained in5. A printed slot antenna with defects in the ground plane that produce five bands was successfully designed in6. Different Euclidean slot forms printed on the ground plane with and without Euclidean patches were used to investigate printed slot antennas7. Two rectangular slots defected on the ground plane and five strips located inside the two slots were employed to operate at four bands  GHz in8.In the past decades, various techniques have been widely employed to design multiband printed antennas based on the defected microstrip structure (DMS), defected ground structure (DGS), meander lines, and notches. These techniques are employed to improve the performance parameters of conventional microstrip/printed antennas for multiband applications. The DGS has been the most widely employed technique in the last two decades. It is implemented by making a defect/etching on the ground plane of microstrip circuits. The defect of DGS can be non/periodic elements or a unit cell9. CMA was applied to design an antenna at 2.4 GHz based on a single meander line in the ground as DGS10. A compact and efficient broadband antenna with two rectangular slots and a reduced ground plane is presented in 11. The performance of this antenna is analyzed using CMT for various radio systems. This paper12 presents a simple compact circularly polarized slot antenna with a wide bandwidth for wireless applications. Its performance is demonstrated using CMA as well as some experimental measurements. A design of a filtering antenna for 5G FR2 band based on CMA of the radiation and feeding structures is presented. Its performance is validated by fabrication and measurements in13. In the study of14, CMT defined the feeding structures and positions for two radiating ground antennas containing simple feeding loops. In15, the CMT was employed to design a compact and wideband circularly polarized antenna predicting the feeding position. CMA was employed to improve the performance of smartwatch antennas16.Characteristic mode analysis (CMA) was recently developed from the characteristic mode theory (CMT). CMA allows a physical insight into the radiation emitted by a multiband antenna and demonstrates how to design an efficient radiating topology that offers a multiband characteristic. The CMT depends on the electric field integral equation based on the method of moment (MoM) formulationIn this work, we present a novel and compact multiband antenna that effectively covers three specific frequency bands relevant to 5G and ITU applications. Our approach incorporates CMA and DGS techniques to facilitate the design and optimization of the antenna. The proposed antenna consists of a defected copper plane with a rectangular slot and three U-shaped slots of different dimensions, creating a DGS that controls the current distributions and the modal behavior of the antenna. The CMA, based on the MoM, is applied to design and optimize the antenna’s characteristics using the equations of CMT. The antenna is simulated using a finite element method (FEM) based electromagnetic simulator and provides three bands with resonance frequencies at 2.96 GHz, 6.06 GHz, and 8.03 GHz. The first and second bands are in the region of 5G sub-7GHz: (2.89–3.03) GHz, and (5.99–6.14) GHz. The third band (7.84–8.21) GHz is appropriate for ITU-8 GHz band communication services. This work demonstrates a novel methodology for multiband antenna design with physical insight, without a substrate, that stands in contrast to conventional antenna design methods that rely on trial-and-error approaches without a comprehensive understanding of the underlying physics. In the subsequent section, we present the key equations derived from CMT that hold particular significance. The proposed antenna undergoes examination using two distinct techniques: firstly, the CMA is employed to achieve the final design of the antenna, following a comprehensive five-step process guided by the equations of CMT. In the subsequent part, the FEM technique is utilized to conduct additional parametric investigations on the proposed antenna. Experimental results are then discussed to validate the simulated outcomes obtained from both the CMA and FEM methods. Finally, a dedicated conclusion section summarizes the findings and implications of the study.17 in 1971 and then investigated by Roger Harington18. It provides the response in terms of eigenvalues, surface current, field radiation, characteristic angle, and modal significance to search for dominant resonating modes. These modes are excitation independent and dimensions dependent. The following eigenvalue equation is used to estimate the characteristic modes of a conducting surface.n is the eigencurrent, λn is the eigenvalue of the nth characteristic mode. The complex impedance Z is defined as Z = R + jX, such that R and X are the Hermitian real and imaginary components of the matrix Z, respectively. The characteristic angle (αn) represents the phase difference between Jn and the electric field En and can be calculated byThe CMT was first proposed by Robert Garbaczn = 0, therefore αn must be equal to 180°; otherwise, the modes are associated with stored energy. Modal significance (MS) presents the normalized current amplitude with a range from 0 to 1 that can be calculated asResonance modes exist when λn = 0), the MS should be equal to 1.At resonance mode (λ3 which corresponds to (0.32 λg × 0.30 λg × 0.01 λg), a dielectric constant (εr = 3), and tangent loss (tan δ = 0.0013). At the top of the substrate, there is only a microstrip line of feeding length (Lf) and width (Wf). Moreover, at the substrate’s bottom, there is a ground plane of the proposed antenna that involves three U-shaped defected slots. The slots are located in a rectangular shape etched with dimensions (13 × 10) mm2. There is a middle vertical strip with a length equal to 8.5 mm and a width (W5 = 0.6 mm), which is under and parallel to a transmission line. The narrow strip evenly divides the slots that consist of two identical U-shaped slots with length (L1 = 10 mm) and width equal to 6.2 mm. With (W4 = 1.5 mm), the arm slot width for the right and left U-shaped slots are kept fixed. The length (L2 = 7.5 mm) and the width (W2 = 7.8 mm) of the inverted third U-shaped are etched. The vertical arms of the inverted U-shaped slot have a width (W3 = 1 mm). All parameters of the proposed printed antenna are shown in Table The proposed printed multiband antenna’s geometry is demonstrated in Fig. g/2 to realize a multiband operation for the printed antenna 19.The slot antenna is commonly employed for wideband and multiband applications aside from designing the printed slot antenna’s slot length, the radiating element must be around λg) is given by:reff is the effective dielectric constant.The guide wavelength . In processing the proposed multiband antenna design, five prototypes are considered as demonstrated in Fig. 2. A rectangular patch is placed in the primary rectangular slot for Antenna 1 with dimensions (10 × 8.5) mm2 to create a U-shaped slot in Antenna 2 as shown in Fig. 2, and an E-shaped slot is formed as Antenna 3 as in Fig. Since there is no substrate material in CMA, the dielectric constant, εn = 0, αn must be equal to 180° indicating that the first two modes are at (5.74 and 5.95) GHz. The existing mode will be according to the transmission line’s position. The eigencurrents’ profiles for Antenna 1 are an explanation for how to design the feeding line , to excite the first or the second mode  GHz as demonstrated in Fig. The first three modes are investigated using software based on MoM to solve the characteristic modes at a specific frequency. Figure re at 5.7 and 5.95The characteristic angles for the first six modes are shown in Fig. The MS is calculated for the three resonance modes by using equation (3), as demonstrated in Fig. 20. Figure A high-frequency structure simulator based on the FEM was employed to optimize the antenna’s performance11) was measured by using a Rohde & Schwarz ZVB 20 Vector Network Analyzer. The top and bottom sides of the fabricated antenna are illustrated in Fig. A novel proposed antenna was designed, fabricated, and measured. The electrical performance parameter (S11) are shown in Fig. 11|≤ − 10 dB.The simulated and measured results of the reflection coefficient , at resonance frequencies  GHz. The resonating areas obtained in Fig. Figure The proposed antenna is compared with published multiband antennas and summarized in Table The simulated and measured radiation patterns of the proposed antenna for an E-plane (Y–Z plane) and H-plane (X–Z plane) are shown in Fig. 2. This fabrication process served as a means to validate the accuracy and reliability of the simulation results. The experimental evaluation of the fabricated antenna revealed its operational capabilities across three distinct frequency bands. Specifically, the antenna operated within the frequency ranges of (2.93–3.07) GHz with a bandwidth of 0.14 GHz, (6.07–6.21) GHz with a bandwidth of 0.14 GHz, and (7.87–8.47) GHz with a bandwidth of 0.6 GHz. The measured results obtained from the experiments demonstrated a favorable agreement with the simulated outcomes. The successful alignment between the measured and simulated results signifies the suitability of the printed antenna for communication service applications within the 5G Sub-7 GHz and ITU-8 GHz bands. These findings highlight the potential practical applications of the designed antenna in meeting the communication demands of these specific frequency ranges.A printed antenna was specifically designed to cater to 5G multiband applications. The antenna design incorporates three U-shaped slots strategically placed as defects within the ground plane, while the transmission line is implemented on the top of the substrate. The performance of the designed antenna is analyzed and simulated through the utilization of CMA based on the method of moments and FEM (finite element method) simulators. The fabricated antenna was constructed on a RO3003 substrate, featuring dimensions of (21 × 20) mm


In this article, a microwave (MW)/millimeter wave (MMW) aperture-sharing antenna is proposed. The antenna is constructed using two orthogonal columns of grounded vias from a 3.5 GHz slot-loaded half-mode substrate-integrated waveguide (HMSIW) antenna. These vias are reused to create two sets of 1 × 4 MMW substrate-integrated dielectric resonator antenna (SIDRA) arrays. With this proposed partial structure reuse strategy, the MW antenna and MMW arrays can be integrated in a shared-aperture manner, improving space utilization and enabling dual-polarized beam steering capability in the MMW band, which is highly desirable for multiple-input multipleoutput (MIMO) applications. The integrated antenna prototype was manufactured and measured for verification. The 3.5 GHz antenna has a relative bandwidth of 3.4% (3.44–3.56 GHz) with a peak antenna gain of 5.34 dBi, and the 28 GHz antenna arrays cover the frequency range of 26.5–29.8 GHz (11.8%) and attain a measured peak antenna gain of 11.0 dBi. Specifically, the 28 GHz antenna arrays can realize dual-polarization and ±45° beam steering capability. The dual-band antenna has a very compact structure, and it is applicable for 5G mobile communication terminals. New spectrum resources have been introduced in 5G to increase data capacity, mainly including the sub-6 GHz (450 MHz–6 GHz) and MMW bands (24.25–52.6 GHz) ,2,3. TheNumerous novel and creative shared-aperture designs using various technologies have been proposed. The first category is the stacked structure ,11,12,13In general, the current shared-aperture antennas still have some flaws: (1) the antenna sizes are large, making them difficult to be applied to the mobile terminals; and (2) most designs cannot support dual-polarized and wide-angle beam scanning in the MMW band.x-polarized and y-polarized arrays, respectively [λ02 (λ02 is the wavelength in free space at 28 GHz) and each element can be fed independently, the MMW arrays can support ±45° beam steering [In light of these issues, this article suggests a 3.5 and 28 GHz aperture-sharing antenna that is based on partial structure reuse strategy. ectively . Since tsteering . To exparε1 = 6.15 and a loss tangent of 0.002, while Sub2 is made of Rogers 4003C with rε2 = 3.55 and tanδ = 0.0027. The coaxial probe numbered #1 excites the 3.5 GHz HMSIW antenna. The inner conductor of the probe is inserted into Via1, while the outer conductor is soldered to the ground. x- and y-directions, and the element interval is 0.51λ02. Two sets of 1 × 4 “H-shaped” slots, which are used to feed the arrays with the microstrip line ports numbered #2~#9, are etched on the ground plane. Proper array spacing and the individual feed of each element enable the wide scanning angles of ±45° for the MMW arrays. All the antenna parameters are listed in the caption of Since the proposed partial structure reuse strategy results in very little interaction between MW and MMW antennas, the two parts can be designed separately.11 mode to resonate at 3.5 GHz for potential 5G applications. 11, TM22, and TM33 modes of the full-mode SIW (FMSIW) resonant cavity at 3.5 GHz, 7.4 GHz, and 8.2 GHz. The electric field in the FMSIW resonant cavity is symmetrical at the central plane along the x-direction in 11|, which reveals three resonant modes in the 3–9 GHz frequency range. 11 mode at 3.5 GHz, TM22 mode at 7.1 GHz, and TM33 mode at 8 GHz, respectively.The 3.5 GHz MW antenna design can be considered first. The preliminary dimension of the antenna can be calculated according to the classical SIW antenna theory ,37 by se11 mode is thoroughly narrow because of the compact size of the antenna. The idea is to shift the higher-order mode TM22 mode downward to combine with the TM11 mode in order to increase its bandwidth. The slot-loading technique is used in the design to implement this idea. 11 mode will not be affected since the current is parallel to the slot, but the TM22 will be significantly affected (shifted down) as the current is cut by the slot (current path will be increased). Similarly, if Slot2 is further introduced, the TM22 mode will continuously be affected because the current is still cut by the slot, and the TM11 mode will also be slightly affected since a small amount of current of this mode is cut as well.The operating bandwidth of the TM22 mode decreases rapidly while the frequency of TM11 roughly remains constant with the increase in length (gL). As can be learned through gW), the frequency of the TM11 mode slightly shifts down and the frequency of TM22 continues to decrease quickly. Finally, the two modes can be merged, broadening the bandwidth from 1.3% to 3.4%.A parametric study of the slot lengths is conducted to clearly verify the validity of the proposed slot-loading technique. With reference to 11| of a single SIDRA element and the gain of a 1 × 4 array. The proposed slot-feed SIDRA structure has two resonances, which are the lower-frequency resonance from the DRA mode and the higher-frequency resonance from the feeding slot mode. The element has a relative impedance bandwidth of 12.5% (26.1–29.6 GHz), while the peak antenna gain of the array is 11.5 dBi. h2 of the substrate, the lower resonant mode shifts down while the higher one stays stable, which indicates that the lower resonance is affected mainly by the DRA mode. sL increases, the higher resonant frequency shifts down while the lower resonant frequency remains almost fixed. This confirms that the higher resonance is influenced mainly by the slot mode. The design of SIDRA has been well presented in . Figure x-axis polarization and y-axis polarization, respectively. Here, a 1 × 4 array along the x-axis is taken as an example to demonstrate the beam scanning characteristic. ψ, 2ψ, and 3ψ are implemented in the elements. The radiation angle θ can be calculated byc is the velocity of light, d is the interval of the elements, which equals 0.51 λ02. The required phase shift ψ to generate the given beam position θ is calculated by Equation (1), and the relative phase shifts of ψ, 2ψ, and 3ψ are assigned to each excitation port [The two sets of 1 × 4 SIDRA arrays can steer the beams along ion port . Figure ion port .r1 > 6) should be chosen due to the need for a compact size of MW antenna and the construction of MMW DRA. Considering the co-aperture of MW and MMW antennas, the thickness of the substrate should be balanced with the planar size of the MMW antenna. Then, several design steps are given as follows:Based on the above analysis, a brief design guideline can be concluded. First, a substrate with high permittivity /2 and λ01 is the wavelength in the vacuum at 3.5 GHz [Determining the initial size of the MW antenna according to the following empirical formula:  3.5 GHz ,37.mL can be set to be around λ02/2.Determining the initial size of the MMW antenna. The initial calculation can be performed with the classical dielectric waveguide model (DWM) for the SIDRA . The sizReplacing the grounding vias of the MW HMSIW antenna with MMW SIDRA arrays and introducing the slot-loading technique to expand the MW impedance bandwidth.Optimizing the final structure to achieve good impedance and radiation performance.The two antennas can be combined to form a shared-aperture antenna once the independent designs of MW and MMW antennas are complete. To testify the coupling between the two different bands and the two polarizations in the MMW band, the S-parameters between different feeding ports should be investigated. As is shown in Z0 = 50 Ω through a gλ impedance converter. The function of a resistor is to improve the isolation of the output port of the power divider, with resistance R = 2 × Z0. The model of the resistor is a 100 Ω Panasonic resistor packaged as 0402.To verify the viability of the presented concept, the co-designed antenna was fabricated as shown in 11|s in the MW band. The measured impedance bandwidth is 3.4%, covering 3.44–3.56 GHz. The measured and simulated peak gains are 5.34 dBi and 5.72 dBi. As described in 11|s for the MMW array are plotted in The measured, simulated gains and |SThis article proposed, simulated, and measured a compact aperture-sharing antenna operating at 3.5 and 28 GHz bands with dual-polarization and beam steering in the MMW frequency band. The slot-loading technology is introduced in the 3.5 GHz HMSIW antenna design to broaden the bandwidth with mode analysis and parametric study conducted. The reuse of the orthogonally arranged grounded vias of the HMSIW antenna enables the dual-polarized beam steering ability for 28 GHz SIDRA arrays. The measurement results indicate that the MW and MMW antennas can work independently. The proposed shared-aperture antenna could be an attractive competitor for wireless communication terminal applications.


As UWB technology develops and devices become smaller, miniaturization techniques for an array antenna system are required. In addition, more in-depth research is needed for UWB direction-finding techniques using channel impulse response (CIR) data. This paper proposes an ultra-wideband (UWB) antenna using a single-radiator multiple-port (SRMP) design for the direction-finding systems of smart devices. The proposed SRMP antenna was designed using a single tripod-shaped patch that can replace the array system. The tripod-shaped radiator was optimized using the edge shape design function to improve its broadband and mutual coupling characteristics. For performance verification, the proposed antenna was fabricated, and the reflection coefficient, mutual coupling, and radiation patterns were measured in a fully anechoic chamber. The proposed antenna has an operating frequency band of 6.1 GHz (from 5.8 GHz to 11.9 GHz) for port 1 and a measured mutual coupling of −14.8 dB at 8 GHz. The SRMP antenna has measured maximum gains of 3.5 dBi for port 1 and 2.9 dBi for port 2. To examine the direction-finding performance, the fabricated antenna was connected to a circuit module with a DW3000 chip, which is widely employed in commercial mobile UWB systems. The direction of arrival (DoA) results using the measured CIR data show root-mean-square (RMS) errors of 1.57° and 4.58° at distances of 30 cm and 60 cm. Recently, there has been a growing interest in ultra-wideband (UWB) localization technology at a short distance for application in indoor real-time location systems, automotive radar systems, virtual reality, and augmented reality ,2,3,4,5.In this paper, we propose a single-radiator multiple-port (SRMP) antenna for UWB direction-finding applications which is more improved with a patch-type design based on the previous publication . The prow1 and w2 and a length of l1, as shown in rε = 4.3, tan δ = 0.018) with a radius of 1r and a height of h. The edge curve of the tripod-shaped radiator is designed with the function f(x) to obtain broadband characteristics while maintaining low mutual coupling. The design function f(x) is the quadratic polynomial expressed in Equations (1) and (2), where k is determined by w1, w2, and l1. The three ports are arranged at the edge of the tripod-shaped radiator at a distance of r2 from the center. The proposed SRMP antenna was modeled and simulated using the CST Studio Suite EM simulator [imulator . A genetzx-plane for ports 1 and 2. The maximum gain of 3.5 dBi is observed at θ = 30° for port 1 and 2.9 dBi at θ = −28° for port 2, within the small aperture size of 30.1 mm. The radiation efficiency and the envelope correlation coefficient (ECC) were also investigated for the proposed antenna. The simulated radiation efficiency of 75.9% is observed at 8 GHz for each port, and the average ECC of 0.134 is obtained between port 1 and port 2 at the target frequency range. f(x) for the upper right corner of the tripod-shaped radiator. The bandwidth characteristics were examined by varying the design function f(x) into an elliptic arc, a circular arc, and the proposed quadratic polynomial as shown in three inset figures. The tripod radiator with f(x) of the elliptic arc (dotted line) does not operate in the target frequency band. The result with f(x) of the circular arc (dashed line) has a broad bandwidth in the high-frequency band over 8.8 GHz. However, it does not achieve a reflection coefficient of less than −10 dB at 8 GHz. By contrast, the tripod-shaped radiator with f(x) of the quadratic polynomial (solid line), shown through Equations (1) and (2), can operate in the entire frequency band from 6 GHz to 12 GHz. Therefore, the essential design parameters of widths 1w and 2w were investigated to optimize the broadband characteristics with low mutual couplings using a single radiator without an additional component. The variations in the bandwidth according to 1w (from 4 mm to 7 mm) and 2w (from 7 mm to 10 mm) are presented, as shown in 2w is wider than 10 mm. 1w and 2w. When 1w is 4.87 mm, the results of the mutual coupling are less than −12 dB. As a result, the f(x) of the tripod structure is optimized with 1w of 10.1 mm and 2w of 4.87 mm. The resulting mutual couplings are less than −14.8 dB, while maintaining a fractional bandwidth of 76.3%. The optimized design parameters of the proposed antenna which are improved with a tripod-shaped radiator using design function f(x) are compared to previous publications as listed in To investigate the DoA estimation performance of the proposed antenna using the CIR data, the DW3000 chip, which is widely employed in commercial UWB direction-finding systems, was used. As the transmitting antenna module, a test board with a monopole chip antenna and the DW3000 provided by Qorvo was used, as shown in ϕ = 0°). The DoA estimation results measured at distances of 30 cm and 60 cm are indicated by the blue and red solid lines, respectively. The results are compared with the ideal result denoted by the black solid line. The square marks represent the average measured values estimated by 20 signals, while the horizontal bars show the measurement error range with the maximum and minimum values, respectively. The root-mean-square (RMS) errors between the mean of the measured results and the ideal value are 1.57° at a distance of 30 cm and 4.58° at a distance of 60 cm. ϕ = 60°), and this measurement was conducted under the same condition. RMS errors of 2.48° and 4.72° are observed at distances of 30 cm and 60 cm, respectively. The results demonstrate that the proposed SRMP antenna using the tripod-shaped radiator is suitable for UWB direction-finding system usage in smart devices.In this paper, we proposed an SRMP patch antenna for UWB direction-finding systems. Considering the limited mounting space of smart devices, the proposed antenna was designed using a single tripod-shaped patch that can replace the array system. The SRMP antenna was optimized using the edge shape design function of the tripod-shaped radiator to improve the broadband and mutual coupling characteristics. The proposed antenna had an operating frequency band of a bandwidth of 6.1 GHz for port 1 (from 5.8 GHz to 11.9 GHz). The simulation of −14.9 dB agreed well with the measurement of −13.3 dB at 8 GHz. In addition, the measured and simulated mutual couplings had similar results of −14.8 dB and −15.5 dB, respectively. The SRMP antenna had measured maximum gains of 3.5 dBi for port 1 and 2.9 dBi for port 2. To investigate the DoA estimation performance of the proposed antenna, the DW3000 chip, which is widely employed in commercial mobile UWB systems, was used. The proposed antenna was connected to the receiving module, and the CIR data were measured at 5° intervals at distances of 30 cm and 60 cm. The DoA results using the measured CIR data had RMS errors of 1.57° and 4.58°. The results demonstrated that the proposed SRMP antenna using a tripod-shaped radiator was suitable for UWB direction-finding system usage in small smart devices.


A high measured efficiency of 92% and a measured gain of 6.52 dB is also achieved. The suggested UWB can effectively cover several wireless applications such as WLAN, WiMAX, and C and X bands.The present study proposes a new, highly efficient fractal antenna with ultra-wideband (UWB) characteristics. The proposed patch offers a wide simulated operating band that reaches 8.3 GHz, a simulated gain that varies between 2.47 and 7.73 dB throughout the operating range, and a high simulated efficiency that comes to 98% due to the modifications made to the antenna geometry. The modifications carried out on the antenna are composed of several stages, a circular ring extracted from a circular antenna in which four rings are integrated and, in each ring, four other rings are integrated with a reduction factor of 3/8. To further improve the adaptation of the antenna, a modification of the shape of the ground plane is carried out. In order to test the simulation results, the prototype of the suggested patch was built and tested. The measurement results validate the suggested dual ultra-wideband antenna design approach, which demonstrates good compliance with the simulation. From the measured results, the suggested antenna with a compact volume of 40 × 24.5 × 1.6 mm Patch antennas have increased in popularity dramatically since the 1970s due to advances in microelectronics technology in the areas of miniaturization and electronic integration. In aeronautical, aerospace, and military contexts, antennas that are compact, lightweight, affordable, highly effective, and easy to install are critical requirements. These low-profile antennas are necessary to fulfill these needs.In contemporary communication systems, printed antennas, which are also referred to as patch antennas or microstrip antennas, are extensively utilized. This is because commercial wireless applications face the same limitations that were once encountered in earlier times due to the proliferation of telecommunications. There exists a plethora of antenna types. In addition, wireless communication devices require more and more frequency bands due to the increasing requirements for wireless services. Since these devices are also intended to be smaller in size for real estate purposes, antennas need to reduce their size and operate in more than one frequency band while maintaining their performance.Multiband and broadband antennas are two types of antennas that are commonly utilized in different communication systems. A multiband antenna is specifically designed to function on multiple frequency bands, making them ideal for wireless communication systems such as cellular networks. These antennas can support various frequency bands, which enables simultaneous transmission of both voice and data. Conversely, broadband antennas are designed to operate over an extensive range of frequencies, utilizing broadband elements like spiral or log-periodic antennas to cover a broad frequency range. This type of antenna is typically used in radar systems that require the detection of signals over a wide frequency range. In conclusion, multiband antennas operate over several frequency bands, while broadband antennas function across a wide frequency range. Both types of antennas offer unique advantages and are utilized in various communication and sensing systems.An antenna for ultra-wideband (UWB) is a type of antenna that is designed to work efficiently over a wide range of frequencies, typically from a few hundred megahertz to several gigahertz. UWB antennas can come in various forms, such as monopoles, dipoles, patch antennas, and horn antennas. Some popular UWB antenna designs include the planar inverted F antenna (PIFA), the tapered slot antenna (TSA), and the printed monopole antenna.To meet the demands of contemporary wireless communication systems, antennas with a low profile, compact size, multiband, and wideband features are greatly desired. Over the past ten years, most modern wire-free communication systems have been developed remarkably swiftly. In order to achieve high-speed broadband connections with minimal power consumption, radio networks use ultra-wideband (UWB) telecommunications technology. UWB was first designed for use with commercial radars. The two primary uses of UWB technology are in consumer electronics and wireless personal area networks (WPANs). Since its early successes in the mid-2000s, wireless UWB technology has emerged as a skill with a small number of smart structures, including the fields of medical engineering, wireless communications, and radar . Owing tDue to its broad bandwidth, high data rate capabilities, power efficiency, interference-free transmissions, effective spectrum usage, secure communication system, and straightforward circuitry for implementation, UWB technology has attracted a lot of interest during the past 10 years . The Fed3 to construct an octagonal UWB fractal antenna. By using an RT5880 substrate and a CPW feed, Niamat et al.  and [6.00–6.60] with bandwidths of 0.2 and 0.60 GHz, respectively. The structure of the antenna has evolved in Iteration 1 to improve the results and fulfill the intended applications. The operating bands of Iteration 1 are considerably improved. However, in this stage, the antenna operates on three impedance bandwidths of 0.49, 0.74, and 0.66 GHz with a gain that reaches 4.97 dB. Thus, in the next iteration, the antenna design is modified by the introduction of four annals. The antenna of Iteration 2 has a dual-band operation with a wide bandwidth that reaches 4.35 GHz. However, the frequency band (2.70–7.05 GHz) centered on 3.14 GHz is not well suited (low loss return). Then, in the penultimate iteration, the geometry of the second iteration was modified by integrating four new rings into each ring. The new shape enables wideband operation from 2.67 to 7.18 GHz as well as a good match with a gain of up to 6.94 dB. However, the antenna presented in this iteration does not meet the X-band requirement. Thus, to further improve the antenna matching and widen the operating band to cover the Band X requirements, the patch is further evolved to Iteration 4, which gives the desired proposed UWB antenna with an increased operating bandwidth that will cover all frequency bands for the planned wireless communication applications. In the simulation of this iteration, the antenna has a UWB characteristic that covers the frequency band (2.70–11.0 GHz) with a gain that exceeds 7.7 dB. Due to this amelioration, the suggested antenna can effectively cover a variety of wireless communication applications, as listed in This section is dedicated to the study of the effects of certain structural parameters through the HFSS simulator. The ideal dimensional specifications of the proposed patch have an impact on the performance of the antenna. The change in these parameters results in significant variations in antenna performance. The appropriate dimensions of the proposed patch can be determined using parametric studies so that it can work effectively with the best attributes.The effect of the Wp on the characteristic of the reflection coefficient, keeping all other dimension parameters unchanged, is illustrated in The ground plane slot position effects on the S11 parameters of the fractal patch antenna are shown in This section examines the variations in the width “Ep” of the ground plane slot. This is another important design factor to see how it affects the reflection coefficient properties of the suggested patch. To properly comprehend how the UWB patch functions, HFSS software was used for antenna design and optimization. After that process, the suggested optimal patch is manufactured and measured to confirm the simulated results. The front and back views of the prototype, as well as the measurement setup in the anechoic chamber, are depicted in The S11 of the prototype patch is measured with the use of the vector network analyzer. The radiation properties study of the suggested patch in terms of peak gain, radiation efficiency, and the radiation pattern is presented in this section. The measurement results are used to confirm the simulation results. The radiation parameters of the patch are realized in an anechoic chamber, as shown in The simulated and measured 2D radiation patterns in the E and H planes at 3.17, 5.82, 7.86, and 9.16 GHz are displayed in Group delay and time domain characteristics are two different measures of an antenna’s performance in the time domain.The analysis of the behavior of an antenna in the time domain is known as the antenna time domain study, and it is crucial to comprehend the transient and dynamic behavior of an antenna ,37,38. TThe study of the antenna time domain plays a critical role in understanding how an antenna behaves when transmitting or receiving signals that change over time. By examining the antenna’s time–domain behavior, engineers can tailor its design to meet specific requirements, such as radar, wireless communication, or satellite communication.Furthermore, time–domain analysis can help detect potential issues with the antenna’s performance, such as impedance mismatches, noise, or interference. This knowledge can then be used to enhance the antenna’s design and performance, resulting in improved signal quality, increased efficiency, and reduced interference.In summary, studying the antenna time domain is a vital component of antenna engineering, as it provides valuable insights into the antenna’s behavior, leading to optimized design and performance for specific applications.In order to demonstrate the time–domain performance of the antenna, a pair of identical antennas are positioned in front of each other, with one serving as the transmitter and the other as the receiver, and their faces directed towards each other. These antennas are positioned at a distance of five times the wavelength of the lowest operating frequency to establish a far-field environment. The time domain response of the proposed antenna is depicted in In other words, the time–domain characteristics describe how an antenna reacts to changes in the input signals over time, while the group delay describes how the antenna affects the phase of the different frequency components of the signal as they pass through it. Group delay is depicted in 3, an impedance bandwidth of 8.3 GHz, and a high gain of 6.25 dB.3. A fractal ring resonator and a ground plane made up of a rectangular part and a half-disk part with a rectangular slot are used to create the UWB operation. The manufactured prototype’s measured results and simulated results match up reasonably well. The proposed antenna operates at measured bandwidths of 7.33 GHz (2.83–10.16 GHz). Additionally, it reports a maximum measured radiation efficiency of roughly 92% and a measured peak gain of 6.52 dB. The developed patch is lightweight, small in size, and has good radiation parameters that enable it to compete in a variety of wireless communication applications, including WLAN, Wi-MAX, Wi-Fi, ITU, C band, and radiolocation, among others.In this paper, a UWB fractal monopole patch is designed and analyzed for wireless communication applications. With the aid of HFSS software, the proposed antenna’s structure was designed and examined. The developed antenna only needs a tiny area of 40 × 24.5 × 1.6 mm