A monopole antenna operated at 2.45 GHz and embedded with artificial magnetic conductor (AMC) for wearable communication systems is investigated in this article. The proposed antenna is composed of a metalized loop radiator with a coplanar waveguide microstrip feedline which is affixed on a cotton fabric material substrate. As well, a cotton-based AMC surface is utilized to eliminate the body’s absorbed radiation and enhance the gain of the antenna. It is composed of 5 × 5 array unit cells etched with I-shaped slots. Using this configuration, simulations show that the specific absorption rate (SAR) level was significantly reduced. Considering flat and rounded body parts, it was found that the SAR values averaged over 10 g at a distance of 1 mm away from the tissues model were only 0.18 W/kg and 0.371 W/kg, respectively. Additionally, the antenna gain was improved up to 7.2 dBi with an average radiation efficiency of 72%. Detailed analysis with experimental measurements of the cotton-based antenna for different operation scenarios is introduced. The measured data show a good correlation with the electromagnetic simulation results. In WBANs systems, wearable antennas are vital components used for communication near the human body6. This challenging role is reflected in the considerations that take when designing such types of antennas. One of these considerations is the influence on the antenna resonance behavior due to the loading effect of the high permittivity body tissue8. On the other hand, in antenna design choosing flexible materials have to be considered to be utilized close to the rounded parts of the human body. Several kinds of wearable antennas based on flexible materials are studied and investigated by researchers such as textile9, flexible substrate10, dielectric resonators11, polyimide12, polydimethylsiloxane13, paper14, and Kapton15. Among these materials, textiles are preferred due to their lightweight and high flexibility in integration with clothing16. However, the implementation process of wearable antennas using textile fabrics as substrates is more difficult compared to the use of conventional substrates17.Nowadays, WBANs are applied in healthcare and medical applications19. In the literature, several techniques have been reported to reduce the body's absorbed radiation and, consequently, minimize the SAR level36. One of the common techniques is using a reflector below the antenna. Different structures have been utilized as reflectors such as high impedance surfaces (HISs)20, electromagnetic bandgap (EBG) structures26, and artificial magnetic conductor (AMC) surfaces36. These structures can increase the antenna gain and help significantly reduce its overall profile compared to the use of a traditional perfect electric conductor (PEC) structure.As wearable antennas operate near the human body, their radiation can cause damage to body tissues. This effect is examined by evaluating the SAR level by considering a specific part of the human body. To reduce the health risks introduced by the wearable antennas, the SAR values should be below the regulated level36. In27, a flexible reconfigurable antenna backed with an AMC surface that worked at 2.4/3.3 GHz was introduced. Considering a human leg model, the evaluated SAR values do not exceed 0.29 W/kg for both operating bands with increasing in the antenna gain by 3.6 and 2.4 dB, respectively. In28, a Yagi–Uda antenna built on a latex substrate and combined with an AMC surface was presented to operate at 2.4 GHz. Single- and double-layered AMC surfaces were used to minimize the peak SAR level to 0.714 W/kg and increase the peak gain up to 1.8 dBi. In30, the performance of a wearable antenna over an AMC surface based on using a stretch conductive fabric was investigated. The design enabled the antenna to cover both WiFi and the 4G long-term evolution (LTE) frequency bands.Among the reported reflector structures, AMC surfaces have been widely used for backing wearable antennas32. The integrated geometrical configuration reduced the SAR value and improved the gain to 0.0721 W/kg and 2.42 dBi, respectively. In33, a flexible AMC surface was used as a reflector. It provides stable performance and reduction in SAR level. In the reported study, the effect of crumpling of the integrated antenna was analyzed. In34, a flexible antenna backed with an AMC ground plan and operates at 2.4 for telemedicine applications is reported. Utilizing the AMC plane provides a 3.7 dB increase in gain, in addition to a 64% reduction in SAR value. A design of compact wearable antennas resonated around 2.65 GHz is reported in35. The backward radiation was reduced using a metasurface recognized as an AMC plane and modeled with a CRLH transmission line operated at negative modes. The peak SAR value of the proposed antenna is 1.25 W/kg for a 5 mm gap from the human body with a real gain of 0.82 dBi. A dual-band 1.57/2.45 GHz wearable antenna with AMC structures is discussed in36. The antenna has a SAR level lower than 0.12 W/kg and a gain value of about 1.9 dBi at the two bands.A textile antenna with AMC surface for WLAN/WBAN applications was reported inThe proposed antenna is fabricated on textile material to achieve lightweight and high flexibility when integrated with clothing.x-axis and the y-axis.The deformation of the integrated antenna was analyzed in free space as well as when it was placed in the vicinity of the human body, indicating its good suitability for operation when bent at both the The integrated antenna has a realized gain of 7.2 dBi with average radiation and total efficiency of 72% and 60%, respectively.The integrated antenna has a low SAR level averaged over 10 g at a distance of 1 mm away from the tissues model where only 0.18 W/kg and 0.371 W/kg, respectively.In this work, a design of a cotton-based wearable antenna over an AMC surface is proposed for 2.45 GHz applications. The integrated antenna adopts cotton fabric as a substrate to ease integration into clothes. In the CST Microwave Studio, the performance and radiation results demonstrated that the antenna provides excellent on-body performance and achieved SAR values below the regulated limit. Detailed discussions on antenna designs with comparative analysis with recent relevant work were presented. Based on the numerical model, the proposed antenna and the AMC surface were fabricated, integrated, and tested. Good agreements between simulated results and measured data were observed. In the end, we can conclude the contributions of the work as,εr = 1.7. A metallic layer is manually attached to the fabric-base substrate to form the antenna radiator and ground plane. As illustrated in the figure, Antenna 1 consists of an L-shaped radiator fed by a 50Ω co-planar waveguide (CPW) line as the first step of the design. With this configuration, a weak resonance around 3.1 GHz is achieved, as shown in Fig. The design steps of the proposed wearable monopole antenna are depicted in Fig. 2. The geometrical configuration of the proposed unit cell and its equivalent circuit are depicted in Fig. g is modeled as Lground, Cd, Cslot, LP, and Cg, respectively38. The optimized lumped element values are displayed in the caption of Fig. Ld and width, Wd of the I-shaped slot on the phase response is illustrated in Fig. Ld, while it shifted down with the increase of the Wd. The optimized values of the Ld and Wd are 13 and 22.3 mm, respectively. The surface current density distribution at 2.45 GHz is examined in Fig. To eliminate the body's absorbed radiation and enhance the antenna gain, the proposed antenna is placed on an AMC surface to reduce the overall profile of the entire structure compared to the use of the PEC surface. Such surfaces operate as inductor–capacitor (L–C) tank circuit at the resonance frequency and achieve HIS. The proposed AMC surface is designed to achieve in-phase reflection at the antenna’s resonance frequency of 2.45 GHz. It was built on a double-compacted layer of cotton fabric with a thickness of 1.8 mm. It consists of 25 square patch unit cells (5 × 5 array) with I-shaped slots, which occupy a whole area of 122.5 × 122.5 mmh = 3, 5, and 7 mm on the antenna performance in terms of |S11| response and peak gain, respectively. The corresponding effects of varying the AMC array size are shown in Fig. In this section, the performance of the single-band wearable antenna over the designed AMC surface is investigated. Two significant parameters that affect the antenna’s performance were subjected to study, the spacing of the AMC surface below the antenna and its array size. Parametric studies were carried out to adopt these parameters. Figure x-axis (in the L direction) and the y-axis (in the W direction). Five different curvature radii along each of the x-axis (Rx) and y-axis (Ry), namely 40, 50, 60, 70, and 80 mm were separately studied. These values are reasonable representations of curvature radii of various rounded positions of the adult human body. The simulated |S11| responses versus frequency for both bending scenarios are displayed in Fig. In practical applications, the wearable antenna is expected to be bent during the operation. To ensure the reliability of the designed antenna for such a scenario, the integrated antenna was subjected to structural bend along both the In this section, the performance of the proposed wearable antenna backed by the designed AMC surface is investigated when considered for operation in the vicinity of human tissues at a distance of 1 mm. To simulate the antenna performance, the Hugo voxel-based body model presented in CST Microwave Suite was used. The Hugo model is an inhomogeneous human model built from 32 tissues. Each tissue has material properties that reflect the anatomical human tissue properties. In this study, the Hugo model allows the determination of the loading effect of the human body on antenna performance and a detailed analysis of the SAR distributions.11| response and radiation characteristics. Figure 11| responses evaluated in free space in comparison to that of body loading for a flat back and rounded arm of radius 50 mm, respectively. The corresponding far-field radiation patterns are illustrated in Fig. The antenna characteristics for flat and rounded body loading were studied. The evaluation was performed in terms of |SRy = 50 mm) when attached to the human arm. The radiation efficiency is almost stable and varies from 70 to 72% for all cases. As well the total efficiency is almost stable and varies around 60% for all cases.Figure 19. The SAR level is expressed as26:ρ is the mass density of the tissue in kg/m3, and E is the total RMS electric field strength in V/m.The SAR level is used to evaluate the amount of RF (radio frequency) energy absorbed by the human body. The Council of the European Union recommended a SAR value of 2 W/kg averaged over 10 g of tissuesThe SAR distribution of the proposed wearable antenna system is evaluated, considering flat (human back) and rounded body (a human arm of radius 50 mm) models. Figure 11| responses of the proposed antenna is given in Fig. Ry = 50 mm, corresponding to the approximate size of an adult human arm. It can be seen from Fig. In order to examine the practical performance of the proposed wearable antenna system, prototypes were fabricated and subjected to measurement. The proposed antenna and the AMC surface were etched on a conductive layer of 0.035 mm thickness glued to a single layer and a double-compacted layer of cotton fabrics, respectively. The prototype performances were measured through an Agilent N9918A vector network analyzer, where a 50-Ω SMA (SubMiniature A) connector was used to feed the antenna. A comparison of simulated and measured |Sx–z plane and y–z plane at the operating frequency of 2.45 GHz is shown in Fig. For free space measurements, the radiation characteristics of the fabricated antenna prototypes for the four different design cases aforementioned before were assessed in an anechoic chamber StarLab 18 as shown in Fig. 11| responses along with the simulated ones are shown in Fig. For real human on-body measurements, a prototype of the antenna backed with the AMC surface was placed close to an adult’s back and arm. The measured |STable εr = 1.7) with a small overall size of 30 mm × 36 mm × 0.9 mm. To mitigate the body coupling effect, an AMC surface was integrated behind the antenna. The structural deformation of the integrated antenna was analyzed in free space as well as when it was placed in the vicinity of the human body, indicating its good suitability for operation when bent at both the x-axis and the y-axis. Further investigation, the SAR evaluation in the Voxel-based human body model considering flat and rounded body parts indicates that the integrated antenna provided average SAR values below the critical rate. The proposed design also features a directional pattern with respect to the on-body with high radiation characteristics, which makes it attractive for potential wearable applications.In this work, a cotton-based wearable antenna convenient to be integrated with clothing was developed for 2.45 GHz wearable applications. The antenna was printed on a 0.9 mm pure cotton fabric (


According to the measurements in the anechoic chamber, antenna gains with the values of 2.3 dBic and 1.1 dBic are obtained for the 870 MHz and 920 MHz, respectively. Finally, the antenna is integrated into a 3U CubeSat that was launched by a Soyuz launch vehicle in September 2020. The terrestrial-to-space communication link was measured, and the antenna performance was confirmed in a real-life scenario.This article presents a multi-band right-hand circularly polarized antenna designed for the Cube Satellite (CubeSat). Based on a quadrifilar structure, the antenna provides circular polarization radiation suitable for satellite communication. Moreover, the antenna is designed and fabricated using two 1.6 mm thickness FR4-Epoxy boards connected by metal pins. In order to improve the robustness, a ceramic spacer is placed in the centerboard, and four screws are added at the corners to fix the antenna to the CubeSat structure. These additional parts reduce antenna damage caused by vibrations in the launch vehicle lift-off stage. The proposal has a dimension of 77 × 77 × 10 mm Typical applications of this system are mentioned, such as communication, Earth observation, and space exploration [The development of the new space industry is disrupting the space market by making satellite fabrication and launch more affordable. Especially miniature satellites, such as the so-called CubeSat , enable loration ,3,4. In In the recent decade, several highly sensitive modulations known as low-power wide-area networks (LP-WAN) have been proposed to enable connectivity from small terminals to space ,6. With A CubeSat includes a communication subsystem for satellite–ground links. The quality of connectivity strongly depends on the antenna system’s performance integrated inside the satellite. Antenna design is a challenge for researchers due to the limitations of size, mass, operating frequency, and required bandwidth. In ,14,15,163 and an axial ratio bandwidth of 18.75%. An antenna is designed on low-loss Rogers substrates and provides an omnidirectional radiation pattern with a maximum gain of 8 dBic.Regarding operating frequency, antennas operating in a band higher than 1 GHz are often preferred due to their small size and ability to support broadband communications. A wideband patch antenna has been reported that can cover 1.6 to 2.7 GHz with a maximum gain of 8.5 dBi . Due to On the other hand, directional antennas have also been extensively studied ,24,25,262 and realized a gain of −1.8 dB. Another study proposed a slot-based antenna that supports the MIMO scheme mentioned in [2. However, this antenna requires a large clearance area, so mounting it on satellites will be pretty tricky. To solve this problem, by analyzing the characteristic mode of the satellite structure, the paper [Differently from the previous examples, sub-GHz bands enable longer transmission distances thanks to the lower path loss and easier obstacle penetration. In article , an omniioned in . Due to he paper  presentshe paper , a propoDesigning the antenna with a compact size to be compatible with CubeSat integration and operation at low frequencies with specific properties in polarization and pattern is a considerable challenge. In , two freLacuna Space is a satellite IoT network provider founded in 2017 . Lacuna This article presents a low-profile, miniature, and multi-band antenna for 3U CubeSat that supports terrestrial ISM bands with right-hand circular polarization. The antenna is designed based on the combination of a broadband power splitter structure and inverted-F antenna elements (IFA) to obtain the directional radiation pattern. In addition, the proposed design provides bandwidth expansion thanks to coupling with a shorted parasitic element. A CubeSat-suitable robust structure is archived using the plastic bumper as the spacing. The antenna is manufactured, mounted on the satellite frame, and then measured in the chamber to validate the concept. Finally, the antenna was integrated within LS3 Lacuna Space CubeSat, launched in September 2020, then commissioned, and used for in-flight experiments in February 2021.As mentioned above, the satellites supporting UHF communication need to cover multiple bands for different geographical areas. When the satellite reaches different areas, it will adjust its transmission and reception frequencies to the bands allowed in those regions, such as 868 MHZ in Europe, 915 MHz in the Americas, and 915 MHz and 923 MHz in Asia. Therefore, antennas used for satellites need to have multi-frequency band features to ensure worldwide continuous operation. In addition, to optimize transmission quality and to reduce the interference caused by ground-based systems, a high gain with circular polarization and a wide beamwidth radiation pattern are required. For these reasons, Lacuna Space decided to use right-hand circular polarization (RHCP) for uplink and downlink communications.2 and 10 mm in height. The radio frequency connector was placed on the bottom layer of the bottom substrate. Based on the literature, a single resonance quadrifilar structure with such a form factor limitation would reach a 30 MHz frequency bandwidth. In order to extend the frequency band operation, a reconfigurable solution based on digital tunable capacitors or pin diodes was considered too risky. In order to provide a passive dual-band antenna, parasitic elements have been extensively used with an inverted F antenna [To integrate the radiation structure on the 1U face of the 3U CubeSat, the maximal dimensions of the structure were 77 × 77 mm antenna . Then, t antenna , two thiInverted F antenna radiating elements are designed using the bottom substrate as a ground plane. The dual-band characteristic is obtained by adding a parasitic element placed in the inner part of the structure and connected to the ground. Several configurations of the driven and parasitic elements were investigated to find out the best topology to maximize frequency bandwidth and gain. Parametric simulations show that the driven element must be placed on the edge of the structure, with the driven element placed at 8mm in the center of the structure. The parasitic element shorting pin is aligned with the feeding element shorting pin. The width and length of the driven and parasitic elements are carefully optimized to maximize the performance in both bands.As shown in Four 8 mm diameter holes are inserted into the top substrate to enable the screwing of the antenna structure to the 3U frame. The M3 metallic screws are also used for grounding purposes on the CubeSat metallic structure. The 3D and top view layouts are shown in Surface currents at 870 MHz and 920 MHz are presented in As described in the previous section, the radiating structure is composed of four independent IFAs with sequential rotation. To radiate right-hand circular polarization, each element must be fed equally with a sequential phase shift of 0ented in . This soIn order to miniaturize the circuit, the proposed feeding network consists of a balun and two hybrid couplers, as shown in tructure .The proposed design is illustrated in Ansys HFSS software is used to simulate the proposed antenna system; then, the design concept could be confirmed. In the next section, two different cases are presented in order to understand the effect of the 3U satellite structure on antenna performance. The case of “without structure” refers to the antenna alone, as shown in The gain achieved on the broad side versus the frequency is presented in Without the CubeSat structure, the realized gain radiation patterns at 870 MHz and 920 MHz are presented in The proposed concept is manufactured and assembled using two FR4 dual-layer substrates (The measured reflection coefficients of the antenna without and with the structure are presented in For the antenna alone, the radiation patterns of the real gain at 870 MHz and 920 MHz are shown in These measurement results validate the performance of the proposed structure for the targeted application.A picture of the antenna integrated within the 3U structure is shown in  antenna  and an eThe proposed structure was validated in space, and the non-deployable passive antenna was demonstrated to be able to cover multiple bands in the UHF frequency with a 3U CubeSat configuration. It should be noted that the antenna’s performance could be further enhanced using a lower-loss substrate. Due to the limited development time, we selected the FR-4 material to benefit from the space qualification and reduced production time. From a 3D electromagnetic solver simulation with a Rogers 4350b substrate (tan A multi-band right-hand circularly polarized antenna integrated into a three-unit CubeSat has been designed, simulated, and measured. A dual substrate solution based on FR4 material has been manufactured to validate the simulation model. After integration into a 3U CubeSat structure, the antenna provides an RHCP configuration with aperture beamwidths of 134 °C and 144 °C at 870 MHz and 920 MHz, respectively. The proposed antenna has been successfully tested in flight with a communication range greater than 1200 km from an LEO CubeSat to a low-power terrestrial receiver.


We designed a mesoscopic dielectric cuboid antenna connected to a flangeless WR-3.4 open-ended waveguide, and the antenna characteristics at 300 GHz were examined through simulations and experiments. Simulations confirmed that the flangeless design eliminated the flange-induced ripples in the radiation pattern, whose shape varied with frequency, and that the antenna operated in the full bandwidth of the WR-3.4 waveguide (220–330 GHz). Prototypes were then fabricated based on the simulation findings. A prototype with an antenna aperture area of 1.5 mm  In particular, the 300 GHz band is being studied and developed actively worldwide. The development of high-gain antennas is important for 300 GHz wireless communication applications because of the proportional increase in free-space propagation loss with the square of frequency. In general, a trade-off exists between the physical aperture area and gain of an antenna; smaller antennas have lower gains. Therefore, achieving high-speed wireless communication in the 300 GHz band for mobile terminals requires a balance between antenna size and gain, so compact high-gain antennas should be designed.Because of its wide frequency bandwidth, the terahertz band is being considered not only for the backhaul and fronthaul of future high-speed, high-capacity mobile communication systems (beyond 5G/6G)13. A step-corrugated horn antenna with dimensions of 3.2 mm 3. Dielectric resonator antennas (DRAs) made of high-dielectric-constant materials for the 300 GHz band have been investigated, and they are drawing attention as alternative on-chip options to planar antennas18, although antennas with resonant structures exhibit narrowband operation around their resonance frequencies. An approach to achieving wideband DRA operation was proposed where multiple higher-order dielectric resonator modes and cavity mode are simultaneously excited; the peak antenna gain was 8.6 dBi at 290 GHz, with a 3-dB gain bandwidth of 55 GHz (270–325 GHz)10. However, even with such frequency bandwidth extension, the potential operating bandwidth limitation due to the resonant structure of DRAs may hinder the complete utilization of the extensive frequency bandwidth of the terahertz band. A dielectric rod waveguide antenna (DRWA) is a surface-wave antenna that operates without using resonance. It has good directivity in the end-fire direction, and its characteristics have been studied analytically and experimentally, including those in the millimeter-wave and terahertz bands30. The maximum gain and minimum half-power beamwidth (HPBW) can be achieved by DRWAs by following Zucker’s design rules27. However, the length of the rod must be several tens of wavelengths to obtain sufficient gain for short-range communication, which requires considerable space when mounting such an antenna on a device. In12, a DRWA with a 20 mm radiation taper was verified; the 29, a DRWA with a 16 dBi gain and Various types of compact antennas operating in the 300 GHz band have been proposed32. A DCA controls the wavefront using a wavelength-scale dielectric cube. Inherent broadband operation is possible because this antenna does not use resonance. We also demonstrated 17.5 Gbps wireless transmission in the 300 GHz band using a DCA sized 1.2 mm 32. However, these DCAs fabricated and demonstrated in the 300 GHz band were designed with waveguide flanges behind them, which caused ripples in the radiation pattern, especially in the E-plane. Moreover, the maximum gain direction varied with the operation frequency, which degraded the frequency characteristics of the antenna gain in a certain radiation direction. This is a critical problem for terahertz wireless communication systems using wide frequency bands based on frequency multiplexing methods, such as orthogonal frequency division multiplexing (OFDM).Recently, we developed and proposed a mesoscopic dielectric cuboid antenna (DCA) and demonstrated its operation experimentallyThis paper investigates the effect of cuboid dimensions on the characteristics of a DCA connected to a flangeless WR-3.4 waveguide . Simulations show that the flangeless design eliminates the ripples in the radiation pattern. The maximum antenna gain is reduced, even though the gain degradation can be recovered by slightly extending the antenna length. We also find an optimal antenna length for a given antenna aperture area. This optimal length resulted in the maximum antenna gain and a radiation pattern with a high rotational symmetry of the main lobe and a low side-lobe level (SLL). Based on the simulation results, we fabricated three DCA prototypes with different dimensions and measured their basic characteristics, namely, their antenna gains, radiation patterns, and c), as shown in Fig. 33. The phase velocity of an electromagnetic wave in a dielectric is slower than that in free space; therefore, the shape of the wavefront can be changed to generate a photonic jet36. This effect enables the conversion of the spherical waves radiated by the waveguide into planar waves to realize high antenna gains.The proposed DCA for the 300 GHz band has a cuboidal radiating part . These results suggest that the waveguide flange significantly affects the radiation characteristics of the DCA and the use of an OEWG can reduce this impact.Fig. In this section, we show the dependence of the gain and radiation pattern of the DCA on its physical dimensions and discuss the radiation efficiency of the DCA based on the simulations. Moreover, we examine the effect of applying a matching taper to the connection between the DCA and the waveguide and optimize this matching taper to realize WR-3.4 full-band operation (220–330 GHz).a and b) and the length of the connecting part (c). Fig. a at various b. We set a and b affect the gain, and an optimal dimension is observed from the perspective of the gain. For example, when a beyond this value decreases the gain instead of further improving it. As the value of b increases, the value of a that gives the maximum gain for a given b also increases, indicating that increasing the DCA gain requires a larger antenna volume. When First, we examine the dependence of the antenna gain and radiation pattern of the DCA on its physical dimensions at 300 GHz. For simplicity, we use a simplified OEWG model, just a straight waveguide with a wall thickness of 0.1 mm. The DCA has three dimensional parameters: the lengths of the cuboid  for a given a. The radiation patterns of both the E- and H-planes have shoulders at approximately b increases from 1.4 mm, the gain increases, and the position of the side lobe in the E-plane pattern becomes closer to the center, resulting in unimodal main-lobe patterns in both the E- and H-planes. At b range, the main lobes in the E- and H-plane patterns are almost identical; i.e., the radiation patterns have high rotational symmetry. In the E- and H-plane patterns, the first side lobes appear at 33 and 35 degrees from the main lobes, with SLLs of b increases from 2.35 mm, the SLLs increase. As shown in Fig. a varies from 1 to 2.2 mm when b is fixed at 2.35 mm. The radiation patterns for the representative dimensions are also shown. For both small  works simply as a waveguide filled with dielectric and does not play a significant role in return loss reduction. Without the matching taper, Finally, we examine the frequency characteristic of We fabricated three prototypes and measured the frequency characteristics of the antenna gain and mentclasspt{minima32 (b, the antenna volume reduced from 1.96 to 1.70 mmDCA1 was designed to obtain approximately the same antenna gain as that in32 with a waveguide flange. As seen in Fig. b characteristic. Based on the simulations, b was set to 1.7 mm for the maximum gain at DCA2 was designed to obtain the maximum antenna gain with the same aperture area (DCA3 was designed to achieve higher gains while maintaining a compact antenna size. We set We measured the antenna characteristics by inserting the DCA directly into the OEWG (Fig. 32, the antenna gain of DCA1 at 300 GHz was 14.6 dBi, which was almost the same as that of the DCA with the waveguide flange. Although DCA2 had the same aperture area as the DCA with the waveguide flange in32, its antenna gain was 16.1 dBi, which was 1.1 dB higher than that of the DCA with a waveguide flange; this antenna gain was obtained at 300 GHz by increasing b from 1.36 to 1.70 mm. DCA3 had an antenna gain of 17.2 dBi at 300 GHz, and the 3-dB bandwidth covered the entire WR-3.4 waveguide bandwidth. The aperture efficiency A is the physical aperture area of the antenna, and G is the antenna gain. DCA3 had a high aperture efficiency of mentclasspt{minimaWe designed a DCA connected to a WR-3.4 OEWG and performed simulations and experiments to confirm their characteristics in the 300 GHz band. For a given DCA aperture dimension, a specific DCA length achieved the maximum antenna gain, a main lobe with good rotational symmetry, and low SLLs. Based on these findings, a DCA prototype with an antenna aperture area of 1.5 mm


The proposed antenna can be easily printed on a piece of flexible tattoo paper and transformed onto a PDMS substrate, making the entire antenna structure conform to the human body for achieving a better user experience. Here, a layer of frequency selective surface (FSS) is inserted in between the antenna and human tissue, which has successfully reduced the loading effects of the tissue, with 13.8 dB improvement on the antenna gain. Also, the operating frequency of the rectenna is not affected much by deformation. To maximize the RF-DC conversion efficiency, a matching loop, a matching stub, and two coupled lines are integrated with the antenna for tuning the rectenna so that a wide bandwidth (~ 24%) can be achieved without the use of any external matching networks. Measurement results show that the proposed rectenna can achieve a maximum conversion efficiency of 59.0% with an input power of 5.75 μW/cm2 and can even exceed 40% for a low input power of 1.0 μW/cm2 with a 20 kΩ resistive load, while many other reported rectennas can only achieve a high PCE at a high power density level, which is not always practical for a wearable antenna.This paper presents a 35.0 × 35.0 × 2.7 mm However, one main limiting factor of most commercialized wearable electronics is their power supplies2. Most of these electronic devices are battery-operated, but unfortunately, the battery itself has limited lifespan and its bulk size does not scale down as fast as electronics3. Nowadays, with the rapid development of 5G technology, where beamforming techniques are massively used, microwave wireless power transfer (WPT) has become an attractive solution to resolve the power charging problem4.Wearable electronics have attracted much interest in recent years due to their wide applications in our daily life. They can be applied on many places such as smartwatches, smart clothing, and real-time health monitoring devices6. Wearable antenna also can be made by electroplating thin metal foils such as copper7 and gold8 on elastic dielectric substrates as well as inkjet-printing conductive nanoparticle inks9 onto flexible substrates. Soft fabrics are chosen for wearable antennas due to their good conformality, flexibility, and low cost10. Despite these excellent features, woven fabric may have a dielectric loss as high as 8.5 dB/m11. Also, gain reduction has been observed in the embroidery antenna reported in Ref.5 due to the higher resistance of the conductive yarn. Inkjet-printed antennas that are generated using the conductive nanoparticle ink on flexible substrates, such as Kapton and PET, are also reported in Refs.12. Despite these conductive nanoparticle inks can provide high conductivity, they can only be printed on specific substrates and carrier mediums13. For example, the self-sintering silver ink reported in Ref.12 can only achieve a low resistance with the use of commercial printing sheets. This will surely limit the compatibility of conductive nanoparticle ink since the radiation efficiency of the antenna is much affected by the dielectric loss of the substrate and the conductivity of the conductive ink trace14. Also, precise deposition of the conductive ink on the substrate requires a complex process, making the fabrication become a slow and non-scalable process15. A wearable antenna that was fabricated using electroplating thin metal foils such as copper and gold on elastic dielectric substrates was also reported in Refs.8. However, these antennas are not able to withstand tensile strain16. Therefore, there is a desire to have a compact, flexible, stable, conformal, and easy-to-make microwave antenna for wearable applications.Flexible microwave antennas that are made of fabric materials have been broadly reported for wearable applications in recent years17, loop antenna18, patch antenna11, and fractal antenna19. Unfortunately, most of the reported rectennas, such as the one with an off-center fed dipole antenna 17, a dual-band rectangular loop antenna 18, and a microstrip patch antenna 11, have a large footprint. In practice, to achieve better user experience, the wearable antennas are required to be compact, low profile, and lightweight20. However, very few of the reported solutions can meet the above-mentioned criteria. Besides that, a rectenna always requires the use of either impedance transforming/matching networks, resistance compression networks, or frequency selective networks, such as those reported in Refs.23, which will surely introduce additional losses and further increase the circuit complexity23. A compact fractal loop antenna reported in Ref.19 was incorporated with an in-loop ground plane (ILGP) so that it can provide sufficient impedance for matching with the rectifying circuit. However, again, it requires the use of a balun and multiple vias for the integration, which has increased the complexity of the circuit design process. Designing a compact and highly efficient 50 Ω rectenna that does not need a matching network remains a challenge for microwave WPT.For microwave WPT, it involves a rectenna that has an antenna to receive the microwave power and a rectifier to convert it to DC power. Different types of rectennas have been reported, including those made of dipole antenna24. In our work, a tattoo-polymer 3 × 3 frequency selective surface (FSS) is integrated with the antenna to isolate it from the lossy human body. Both the antenna and FSS here can be rapidly and easily fabricated using the commercial tattoo paper and liquid metal alloy. The “Ag-In-Ga” traces in our antenna structure are coated with silver epoxy and liquid metal for achieving a higher conductivity (3.8 × 106 S/cm). Our method has provided an alternative solution for the crucial problem frequently encountered by most fabric antennas, whose conductivities are usually low due to the limitation posed by embroidery density5. Different from the state-of-the-art rectennas23, here, a matching loop, a matching stub, and two coupled lines are employed for tuning the antenna impedance so that it can achieve a broad-band impedance matching with the rectifying circuit. The complexity of the antenna structure is simple as it does not require the use of any external matching networks. Finally, as our antenna is having 50 Ω impedance, it can integrate easily with the commercialized rectifiers, such as that demonstrated in Ref.25.In this paper, a compact, conformal, and easy-to-make tattoo-polymer loop antenna is presented for designing a highly efficient rectenna. As human body is very lossy, it can cause the antenna’s radiation performance to deteriorate significantly. Degradation in the radiation pattern can result in transmission errors3, as schematized in Fig. 26 of all the layers are given in Supplementary Table 14. Finally, the tattoo antenna and the FSS were transferred to the PDMS substrate  through the hydroprinting process. The PDMS is selected to be the structural support of the antenna due to its higher flexibility, lower cost, and easier fabrication, as compared with other polymer substrates27. The total profile of the proposed tattoo-polymer antenna is 2.70 mm, which is thinner than most of the state-of-the-art EBG/FSS-backed wearable antennas, as summarized in Supplementary Table S232. Description of the detailed fabrication processes can be found in Refs.33. The final designs of the antenna and FSS are shown in Fig. In order to simulate the proposed tattoo-polymer antenna, we have employed a multilayer tissue model (Syndaver) comprising skin, fat, and muscle layers with an overall dimension of 20 × 9 × 11 mm19, it is usually very challenging to miniaturize its footprint as the circumference of a typical loop resonator is required to be in the multiple of its operating wavelength. Our proposed antenna is employed for designing a compact rectenna that can operate in the Wi-Fi range (2.40–2.48 GHz). Here, the octagonal loop is more preferred than the conventional square loop. This is because the 90° sharp bends of the square loop tend to generate stray radiation, and it can degrade the performance of the antenna34. To lower down its resonant frequency, here, two coupled lines are loaded to the loop antenna. A matching loop is appended to the loop for optimizing the antenna impedance, which will be discussed in detail shortly. The design procedure starts with simulating the reflection coefficient of a simple octagonal loop without the two coupled lines in free space. With reference to Fig. λ = 96.15 mm), which corresponds well to the perimeter of one circumference (101.15 mm). With the inclusion of the two coupled lines (L6 = 11.00 mm), as can be seen from the same figure, the resonant frequency has shifted down to 2.94 GHz. After performing the simulation in free space, which can be done in a faster manner, the antenna is now attached to the human tissue model for further optimization. Due to the loading effect of the high-dielectric lossy tissue35, the resonant frequency of the antenna has further shifted down to 1.86 GHz, which has caused poor impedance matching. It shows that the tissue layer has deteriorated the impedance performance of the antenna significantly due to its high loss. Then, a 3 × 3 FSS layer is inserted in between the loop resonator and the human tissue as an isolator. With the inclusion of the FSS and PDMS substrates, the antenna is fine-tuned so that it operates at 2.40 GHz with good impedance matching, as shown in Fig. λ2.4GHz) in Ref.18 and the fractal loop  in Ref.19, the circumference of our octagonal loop antenna  is 21.8% and 56.1% smaller than the former and the latter, indicating that ours is more compact.Although the loop antenna has been employed for harvesting energy36. To solve this problem, our antenna is integrated with a matching loop, a matching stub, and two coupled lines for tuning the antenna impedance for matching with the 50 Ω rectifier. By tuning the lengths of the matching loop (M1) and the matching stub (L7), good impedance matching can be easily realized over a wide bandwidth, as shown in Fig. Since the antenna is to be used for WPT, it is very important for the tattoo-polymer antenna to have a good impedance matching with the RF-to-DC rectifier circuit for maximizing the RF power transfer. Although matching circuits and filtering networks can be introduced for improving the impedance matching, they can increase the complexity and size of the antenna system37, the ring FSS has low back radiation when it is integrated with a dipole. First, the FSS element is placed on the human tissue model and simulated inside a unit cell for finite structure, which is available in the CST MWS software, as shown in Fig. z and − z directions. With reference to the same figure, two magnetic walls (Ht = 0) are set in perpendicular to the x direction, while two electric walls (Et = 0) are vertically defined in the y direction. Figure R = 5 mm (0.042λ2.4GHz), 4 mm (0.032λ2.4GHz), and 3 mm (0.025λ2.4GHz)] in the phase range of 180° to − 180° with respect to frequency. The reflection phase is obtained by de-embedding the reference plane to the surface of the element. A reflection phase of ~ 0° is observed near to the operating frequency of 2.40 GHz38 for the case R = 0.025λ2.4GHz. This is to make sure that the reflected wave is in phase with the incident wave on the tissue model39. The corresponding transmission and reflection coefficients are also simulated and shown in Fig. R = 0.025λ2.4GHz, the reflection and transmission coefficients are found to be − 30.7 dB and − 7.45 dB, respectively, which correspond to 0.029 and 0.424 at the operating frequency of 2.40 GHz. For the calculated insertion loss 1 − |S11|2 − |S22|2 = 0.82, it is observed that the human tissue is extremely lossy due to its high insertion loss.In this section, the design procedure of the FSS is discussed. The ring-shaped FSS is selected because it has a simple structure. According to a study presented in Ref.35. Here, the SAR simulation is conducted by establishing a multilayer biological human tissue model using the CST software. Then, the tattoo-polymer antenna is placed right above the biological tissue with a separation distance of 5 mm28. The input power to the antenna is set to be 100 mW at 2.40 GHz, and the SAR is calculated according to the IEEE C 95.1 standards. The SAR values are simulated by averaging over the sampled volumes of 1 g and 10 g. By comparing Fig. 40 and the EBG-backed planar inverted-F antenna in Ref.41, the SARs of our FSS-integrated octagonal loop antenna are 64.0% and 27.6% lower than those in Refs.41 by referencing the volume of 1 g tissue sample (USA Standards); while they are 34% and 42% lower than the reported wearable antennas in Refs.42 with 10 g tissue sample volume (European Standards), indicating that our FSS-integrated octagonal loop antenna has achieved a lower SAR with a more compact size. The details of the comparison are summarized in Supplementary Table Since the proposed tattoo-polymer antenna is designed for wearable applications, the specific absorption rate (SAR) is an important factor to be considered as it is required to be placed near to human body43, as depicted in Fig. 26. Figure The tattoo-polymer antenna was integrated with the FSS and fabricated, and experiment was carried out using a differential probeyz-plane, xz-plane, and xy-plane) of the integrated tattoo-polymer at 2.40 GHz when it is placed on the artificial skin layer. For comparison purpose, the radiation patterns of the loop antenna on tissue without FSS . A considerable gain improvement (13.8 dB) in the boresight can be achieved by integrating the loop antenna with the FSS structure. It shows that the FSS layer has successfully isolated the antenna from the skin, and it has improved the radiation performance significantly.Supplementary Figure 38. Therefore, it is crucial to ensure that the operating frequency of a wearable antenna remains stable even when it undergoes deformation. An experiment was performed based on the structure shown in Fig. In practical applications, a wearable antenna needs to be worn on the human body, which may result in bending or deformation. The resonant frequency and impedance matching of the antenna are susceptible to changes due to structural deformation11|) was measured in Fig. The impact of the backing objects on the antenna performance was also investigated in practical scenarios. The FSS-integrated tattoo-polymer antenna was placed on hand, chest, and leg of a male volunteer as depicted in Fig. S  at 2.45 GHz, using the experimental setup depicted in Fig DC over a 20 kΩ load. It is observed that the output voltage of the proposed antenna is linear, and the optimal load does not vary with different power densities. It implies that the proposed design does not require the maximum power point tracking (MPPT) capability to maintain its maximum PCE11, which is a very desirable feature. After selecting the optimal load (20 kΩ), a frequency sweep is performed from 1.0 and 3.0 GHz at different power densities (S) for measuring the bandwidth of the antenna. Figure S. Figure .1 < S < 5.5 μW/cm2 with the optimal load (20 kΩ). It is observed that the PCE of the rectenna exceeds 40% from 1.0 μW/cm2 onwards, indicating that it is suitable to be used for low power levels as well. As proof of concept, a commercialized digital watch was connected to the output of the proposed rectenna, as shown in Figs. The wireless power performance of the proposed tattoo-polymer antenna was evaluated by conducting a load sweep from 0 to 100 kΩ at three fixed power densities, 44. This is because our antenna does not need the use of external matching circuits and long transmission lines, which can introduce additional insertion loss. With reference to the table, similar level of PCE was achievable for the wearable rectennas in Refs.19. A resistance compress matching network was employed in Ref.11 so that a higher DC voltage sensitivity and a better PCE could be achieved. While in Ref.19, an in-loop ground plane (ILGP) was engaged to adjust a loop antenna’s impedance to ⁓ 50 Ω for attaining a high PCE (61%). The ILGP structure was connected to the loop antenna through a metallic via. For both, however, the employment of these additional external circuits will surely increase the antenna complexity. In contrast, our proposed antenna structure is rather simple as it does not require the use of any external matching networks. Besides that, many other rectennas, which are proposed for WPT and energy harvesting (EH), have put emphasis on maximizing the PCE. However, high PCE was often achievable only at high power density levels (more than 100 μW/cm2)45. For example, Falkenstein et al. reported a dual linearly polarized patch antenna with total efficiencies over 50% for a high power density of 25–200 μW/cm246. Harouni et al. reported a 2.45 GHz compact dual-circularly polarized rectenna with PCE = 63% for a high power density of 525 μW/cm247. Also, high PCEs of 84.4% and 82.7% at 2.45 and 5.80 GHz, respectively, were achieved by the CPS dipole dual-frequency antenna reported by Suh48. Although the reported rectennas have reasonably better PCE at a higher power density, practically, the RF/microwave power available in an ambient environment is usually very low. For example, based on the London RF survey measurement reported by Pinuela49, the average power density of the Wi-Fi (2.40–2.50 GHz) is only 0.89 nW/cm2 in most of the underground stations. The power density was found to be in the range of 0.0017–0.8594 μW/cm2 for the European Union (EU) telecommunication spectrum of 10 MHz–6 GHz under long-term RF EMF measurements50. Therefore, it is important that a rectenna can operate and maintain a high efficiency under an ultralow power density. By comparing with the reported rectennas in Refs.50, our proposed rectenna can achieve higher efficiencies of 8.5% and 7.5%, respectively, with lower power densities of 125% and 110%, indicating that our proposed rectenna can achieve a higher sensitivity.The proposed rectenna is compared with the state-of-the-art low-power rectennas, as shown in Supplementary Table 51, an organic pulse oximetry sensor (minimum power consumption = 24.0 μW)52, and a biomedical electrocardiogram (minimum power consumption = 75.0 nW)53. Based on the results shown in Fig. 4. Although the prospect is promising, the application of the tattoo-polymer antenna to microwave WPT is still in the nascent stages. Many crucial issues, such as wear resistance and long-term durability, are yet to be explored. Further improvements on the rectenna’s maximum output power and PCE are always needed to power up a wider range of electronic devices with a power consumption in the range of mW. Therefore, future efforts will focus on further characterization of the device durability and WPT output power of the proposed tattoo-polymer antenna, which includes encapsulation with a Tegaderm film to improve the wear resistance as well as enhancement in the output power through a commercialized booster module or array unit.In summary, this work has presented the design and measurement of a compact tattoo-polymer FSS-integrated loop antenna for on-body wireless power transfer with high PCE, which is operating well under low power levels, and it is intrinsically matched to 50 Ω without needing any external matching circuits. Proof of concept demonstration shows that WPT can be achieved through the proposed rectenna by harvesting energies from the microwave/RF sources such as the Wi-Fi router and mobile hotspot. Recent advances in the ultralow-power electronics have led to several reported wearable devices with power consumption as low as nW to μW, such as a pressure sensor (minimum power consumption = 10.0 μW)2. The efficiency is found to be able to exceed 40% even for a low input power of 1.0 μW/cm2. It shows that our proposed rectenna can operate well under low power levels.A wearable tattoo-polymer FSS-integrated loop antenna has been proposed for wireless power transfer. Both the antenna and FSS can be easily printed on a piece of tattoo paper. The footprint and profile of the proposed antenna are smaller than most of the state-of-the-art wearable antennas. The FSS layer has successfully improved the radiation performance of the loop antenna on human body in the boresight, with a significant gain improvement of ~ 13.8 dB. Also, the FSS can improve the SAR as much as 75.5% on the biological tissue. Bending test has been performed on the proposed wearable antenna, where the resonant frequency and impedance bandwidth are found to be very stable. The antenna performance is not affected much by its own structural deformation. For wireless power transfer, the FSS-integrated antenna was bonded with a 50 Ω 6-stage voltage multiplier. It has been observed from experiments that a maximum efficiency of 59.0% is achievable with an input power of 5.75 μW/cm28. The effects of backing loading objects are analyzed for realistic scenarios. The FSS-integrated tattoo-polymer was placed on the hand, chest, and leg, as shown in Fig. The proposed FSS-integrated tattoo-polymer was placed on a man-made curvature created by wrapping artificial skin around cylindrical polystyrene foam of diameters 70 mm, 100 mm, and 120 mm to produce various degrees of structural deformation as depicted in Fig. 44 to match the antenna impedance of 50 Ω. Fig. Aeff)47, as indicated by Eq.  was utilized to transmit a continuous wave at 2.45 GHz (with power ranging from 0 to 30 dBm) using a horn antenna as the source. The tattoo-polymer antenna, placed 1.80 m away from the source, was connected to a variable resistive load through its DC output. To meet the minimum Fraunhofer far-field criteria, a distance of 1.20 m was maintained between the horn antenna and the proposed antenna. A 6-stage voltage multiplier, designed using BAT15-03W diodes (low barrier Schottky diodes), was utilized to convert the harvested microwave energy into a DC voltage. The rectifier circuit, fabricated on a flexible PCB, was modified from Ref.Supplementary Information.


It also has the highest gain  compared to other antennas in the literature. In addition, the SAR values for the proposed antenna are well below the safety limits prescribed by IEEE Std C95.1-1999, with SAR values of 0.409 W/Kg for 0.8 GHz, 0.534 W/Kg for 1.43 GHz, 0.529 W/Kg for 3.5 GHz, and 0.665 W/Kg for 5.5 GHz when the applied input power is 10 mW. Overall, the proposed antenna in this study demonstrates superior performance compared to existing tri-band implantable antennas in terms of size, bandwidth, gain, and SAR values.This study aims to design a compact antenna structure suitable for implantable devices, with a broad frequency range covering various bands such as the Industrial Scientific and Medical band , the Wireless Medical Telemetry Service (WMTS) band, a subset of the unlicensed 3.5–4.5 GHz ultra-wideband (UWB) that is free of interference, and various Wi-Fi spectra . The antenna supports both low and high frequencies for efficient data transfer and is compatible with various communication technologies. The antenna features an asynchronous-meandered radiator, a parasitic patch, and an open-ended square ring-shaped ground plane. The antenna is deployed deep inside the muscle layer of a rectangular phantom below the skin and fat layer at a depth of 7 mm for numerical simulation. Furthermore, the antenna is deployed in a cylindrical phantom and bent to check the suitability for different organs. A prototype of the antenna is created, and its reflection coefficient and radiation patterns are measured in fresh pork tissue. The proposed antenna is considered a suitable candidate for implantable technology compared to other designs reported in the literature. It can be observed that the proposed antenna in this study has the smallest volume (75 mm Due to the rapid development of low-power, miniaturized electronic devices and sensors, there is growing interest in implantable medical devices (IMDs) among researchers. IMDs have potential applications in neural stimulation, therapeutic medication, diagnosis and treatment of ailments, deep body communication, and drug delivery devices with high accuracy ,5,6,7,8.Antenna design and performance enhancement techniques are receiving significant research interest due to the numerous factors and challenges involved in designing antennas for implantable devices as compared to free space antennas. The performance of implantable antennas is affected by the behavior of electromagnetic radiation in the highly diverse electrical properties of human tissues, which have heterogeneous permittivity and conductivity based on water content. The major challenges include the coupling of electromagnetic radiation with body tissues due to high conductivity, dielectric constant, and heterogeneous tissue layers, resulting in frequency and impedance detuning effect, signal degradation, poor gain and radiation efficiency, and tissue heating ,10. As aLimited space for integrated circuits and antenna structures designed for implants is another issue. Consequently, antenna electrical dimensions are inversely proportional to resonance frequency and other radiation characteristics. Therefore, active techniques are needed to miniaturize the antenna while maintaining performance in practical in-body applications ,12. ReseNarrowband antennas are not ideal for implantable devices because their resonance frequency is affected by the varying electromagnetic properties of human tissues . WidebanAccording to the above discussion of the most related works with this study, the main contribution of this study is the design of a triple-band implantable antenna with wide bandwidth performance that operates at ISM, WMTS, UWB, and Wi-Fi frequencies, making it suitable for commercial applications. The design incorporates a parasitic resonator and an asynchronous-meandered radiator with an open-ended square ring defected ground. The design is analyzed using a simplified three-layered human tissue model in CST Microwave Studio Suite. A prototype is fabricated and tested by implanting it into pork tissue, and numerical simulations are used to calculate the specific absorption rate (SAR) to determine the allowable input power. The proposed structure has incorporated the following advantages over the state-of-the-art literature.The proposed structure is a compact antenna with simple and planar configurations. Existing structures are designed using multiple layers, shoring pins, and PIFA technology, making the antenna thicker and more complex.Most Tri-band antennas are designed to resonate at ISM and WMTS bands, whereas the literature shows that the Ultra-wideband spectrum has significant advantages for Intra-body communication. Triple-band resonance was selected to make the antenna suitable for commercial body area network applications.The antenna has flexible characteristics, robust to structural deformation, size and shape of the tissue.Wider impedance bandwidth to sustain the frequency detuning effect caused by the heterogeneous body tissue effect in real-time scenarios.Broadside radiation characteristics with better gain for reliable communication link and low specific absorption rates.The paper is structured as follows: rε = 38 and σ = 1.46 s/m), fat (rε = 5.2 and σ = 0.10 s/m) and muscle (rε = 52.7 and σ = 1.8 s/m) layers is designed. According to the in-body deployment location, it may be required to bend the structure for the specific tissue. Thus, antenna performance is also observed for bending. Numerical modelling of tissue: As the proposed structure is designed for implantable application, and a numerical model for the heterogeneous body tissue layers is created. It has three layers: the innermost layer is muscle tissue with a thickness of 20 mm, the middle layer is fat tissue with a 5 mm thickness, and the top layer is skin tissue with a 2 mm thickness. The human body can have different thicknesses of tissues depending upon body type, body fat or mass index, and different organs. In our model, average values for tissue thickness are considered ,37. Elecrε value of 10.2 and δ = 0.0022. The planar size of the structure is 10 mm × 10 mm, which is equivalent to (0.029 λo × 0.029 λo), where λo represents the free-space wavelength at 0.868 GHz (lowest resonance frequency). To use the antenna inside the body tissue, the radiator and ground are covered with the dielectric material layers (RO 3010). It insulates the conducting part of the antenna from lossy tissue layers and also contributes to lowering the resonance frequency. Design evaluation of the proposed structure for attaining wider bandwidth is explained in three steps. Antenna Geometry and Design Process: Configuration and geometry of the designed tri-band patch antenna is depicted in To facilitate a better understanding of how to determine the structure of antennas, we will introduce a mathematical model. This model incorporates Equations (1)–(3), which represent the fundamental free space formulas. Furthermore, to analyze the impact of bio tissues, we have combined these equations with Equations (4)–(7). The details are as follows:rf), resulting in an approximate value of 0.843 GHz, close to the simulated resonance frequency.To achieve a lower frequency of 0.86 GHz, we initially utilized a meandered patch to miniaturize the antenna. The meandered path was incorporated to increase the length of the radiator, thereby extending the electric current path and reducing the resonance frequency . Equatioc represents the speed of light, which is approximately 3 × 108 m/s. The length of the resonating path, denoted as W is the antenna dimension of meandered path and values are listed in In the aforementioned equations, the symbol Considering that the operating environment for the antenna is a complex heterogeneous medium, the dielectric constant of the antenna varies within the layered bio tissue phantom. To account for this, we calculate the equivalent dielectric constant of the antenna, which incorporates the substrate and bio layers. This calculation is carried out using Equations (4)–(7).rε1, rε2, rε3 and rε4. Additionally, the parameters d1, d2, d3 and d4 represent the distances between the boundaries of each layer. These distances can be calculated using the methods described in references [n = 1, 2…4, representing number of substrate layers; w15 is the feed line width, w is the width of the antenna, and The relative permittivity of the antenna substrate and the tissue layers surrounding the antenna substrate are denoted as ferences ,40.(5)dS11| plot for the design steps is represented in S11| above 10 dB indicates the good impedance matching of the antenna, which makes the antenna efficient in terms of radiating the maximum power. S11 = 0 shows no power is radiated from the antenna. In the proposed design, S11 is above 0 at the resonance bands, making the antenna suitable in terms of radiating. To justify the step-wise impedance matching, the VSWR plot is shown in S11|, and VSWR.Further, in step1, an open-end slotted ground is used to improve the impedance matching. Thus, with the coupling between multiple slots of an asynchronous-meandered radiator and open-ended square loop-type ground, the multiple resonance bands are achieved in step1. The |s11| is above 10 due to good impedance matching, making the VSWR value less than two. It is studied from the literature that wider impedance bandwidth can be achieved by combining the resonance modes [fo is the fundamental resonance frequency at 0.8 GHz. In step1, the resonating modes are excited at fo = 0.8 GHz, 2fo = 1.6 GHz, 4fo =3.2 GHz, 5fo = 4.0 GHz, and 7fo = 5.6 GHz. Higher modes (4fo to 7fo) are in close proximity to each other and merged. Thus, wider impedance bandwidth from 3.03 GHz to 7.06 GHz is achieved.In the first step, three resonance frequencies with bandwidths of 130 MHz (0.81–0.94 GHz), 190 MHz (1.56–1.75 GHz) and 4.03 GHz (3.03–7.06 GHz) were attained. Over the resonance bandwidth |ce modes ,42,43. SIt can be found that the reflection coefficient values of the lowest and middle band are 15 dB and 17 dB, respectively. Also, as middle resonance band is not at the desired frequency (1427 MHz to 1432 MHz) for the WMTS band. According to the in-body application, it has been studied in the literature that antenna impedance reduces and frequency may detune due to complex multi-layered body tissue structures. Electrical properties and depth of different tissues vary from person to person as well as for different organs. Therefore, to avoid this issue in practical situations and to tune the middle resonance frequency, the structure is modified in step 2.In the second step, the length of the radiator is increased by adding a three-slot meandered element with the radiator to attain the resonance at the WMTS band. This modification has a significant impact on antenna performance. The middle band is tuned at desired range as well as the reflection coefficient value for the lower and middle bands also improved and shifted to 36 dB and 34 dB. It ensures the improvement of impedance matching. However, the upper UWB spectrum is not affected by this modification. Surface current distribution for step1 and step2 at 0.88 GHz and 1.46 GHz is shown in fo is excited at 2.7 GHz and merged with the resonance mode at 4fo; it widens the bandwidth toward a lower cut-off frequency of the upper band (|S11| plot of step3). In this way, the combination of various modes has contributed to the impedance bandwidth from 2.6 GHz to 6.3 GHz. The length of the parasitic resonator is optimized from both edges. Parametric analysis of the reflection coefficient for the lengths of the left and right edges is shown in In the third step, an inverted U-shaped parasitic resonator is placed around the radiator. Due to capacitive coupling between the radiator and parasitic resonator, the mode at 3S11| parameter above 0 dB in The significance of impedance matching by adding the parasitic resonator is shown in Consequently, a compact triple-band antenna via the embedment of open-ended square ring ground, parasitic resonator and meandered patch establishes the required features for biotelemetry systems.Furthermore, the surface current plot for various frequencies of the antenna is shown in To ensure the accuracy of the simulated results, an antenna prototype was fabricated, and its performance parameters were measured. The antenna was inserted into animal tissue for implant application, and images of the prototype and measurement setup can be found in In the numerical simulation, the antenna is placed within the muscle layer and is effectively isolated from the surrounding tissue by the dielectric layers positioned above and below it. Among the skin and fat tissues, the muscle layer possesses the maximum implant depth, highest dielectric constant (52.7), and conductivity (1.74 S/m). As a result, the muscle tissue has the most significant influence on the antenna’s performance. To achieve a good agreement between the simulated and measured results, pork was employed as it exhibits tissue-equivalent properties. This choice of material enhances the alignment between the simulated and measured outcomes. It is worth noting that variations in simulation and measurement setups have been observed in the existing literature ,14,22,26x-axis, as shown in In order to analyze the stability of antenna performance against the shape of the implant tissue, the antenna is deployed in the skin layer of the cylindrical phantom and numerically simulated; further, the antenna is also bent along the For all of the operating conditions, the antenna has covered the bandwidth for various valuable communication standards, including ISM bands , WMTS band (1427–1432 MHz), and interference-free subset of the unlicensed 3.5–4.5 GHz ultra-wideband (UWB), as well as Wi-Fi frequencies at 3.6/4.9/5/5.9/6 GHz. These findings demonstrate the potential of the antenna for a wide range of communication applications.The performance of an antenna designed for in-body communication must be carefully evaluated to ensure reliable communication links and avoid any harmful effects on body tissues. One key factor in determining the effectiveness of such an antenna is its radiation pattern. To this end, When the antenna is placed below the skin layer in the cylindrical phantom and bent across the radius of 30 mm, both the gain and efficiency reduce. The middle band has the maximum deviation of 1.67% in efficiency and 9 dB in gain. These values provide important information for determining the overall performance of the antenna and its suitability for various communication applications.In sum, the proposed antenna exhibits a radiation pattern that is well-suited for in-body communication applications and is validated through both experimental and numerical simulations. The gain and radiation efficiency plots provide further evidence of the antenna’s effectiveness across different frequencies. These findings represent a significant contribution to the development of implantable devices and pave the way for further advancements in this field.The specific absorption rate is a safety parameter used to measure the amount of heat absorbed by the tissue. The IEEE Std C95.1-1999 sets a SAR limit of 1.6 W/Kg over 1 g of tissue. x-axis at the radius of 30 mm. This analysis confirms the reliability of using the antenna inside different body organs. The antenna shows stable resonance frequencies with a slight reductions in bandwidth at the upper band. Gain and efficiency parameters are also stable over the entire bandwidth.3) and widest bandwidth . It also has the highest gain  compared to other antennas in the literature. In addition, the SAR values for the proposed antenna are well below the safety limits prescribed by IEEE Std C95.1-1999, with SAR values of 0.409 W/Kg for 0.8 GHz, 0.534 W/Kg for 1.43 GHz, 0.529 W/Kg for 3.5 GHz, and 0.665 W/Kg for 5.5 GHz when the applied input power is 10 mW. Overall, the proposed antenna in this study demonstrates superior performance compared to existing tri-band implantable antennas in terms of size, bandwidth, gain, and SAR values.Finally, 3 and the widest bandwidth of 181.8% for 0.86 GHz, 9.58% for 1.43 GHz, and 285.7% for the UWB subset and Wi-Fi. It also has the highest gain, with values of −26 dBi for ISM, −14 dBi for WMTS, and −14.2 dBi for the UWB subset and Wi-Fi. Furthermore, the SAR values of the proposed antenna are well within the safety limits prescribed by IEEE Std C95.1-1999, with SAR values of 0.409 W/Kg for 0.8 GHz, 0.534 W/Kg for 1.43 GHz, 0.529 W/Kg for 3.5 GHz, and 0.665 W/Kg for 5.5 GHz when the applied input power is 10 mW. Overall, this antenna demonstrates superior performance in terms of size, bandwidth, gain, structural deformation, and SAR values, making it a promising candidate for implantable medical devices.This study presented a compact and efficient implantable antenna operating at triple bands. The proposed antenna employs a combination of a multi-open-end slotted meandered radiator, a parasitic patch, and a square ring-shaped ground to excite multiple resonant modes for various biotelemetry applications. The proposed antenna in this study has several notable advantages over other antennas in the literature. It has the smallest volume of 75 mm


Millimeter wave (mm-Wave) wireless communication systems require high gain antennas to overcome path loss effects and thereby enhance system coverage. This paper presents the design and analysis of an antenna array for high gain performance of future mm-wave 5G communication systems. The proposed antenna is based on planar microstrip technology and fabricated on 0.254 mm thick dielectric substrate (Rogers-5880) having a relative permittivity of 2.2 and loss tangent of 0.0009. The single radiating element used to construct the antenna array is a microstrip patch that has a configuration resembling a two-pronged fork. The single radiator has a realized gain of 7.6 dBi. To achieve the gain required by 5G base stations, a 64-element array antenna design is proposed which has a bore side gain of 21.2 dBi at 37.2 GHz. The 8 × 8, 8 × 16, and 8 × 32 antenna array designs described here were simulated and optimized using CST Microwave Studio, which is a 3D full-wave electromagnetic solver. The overall characteristics of the array in terms of reflection-coefficient and radiation patterns makes the proposed design suitable for mm-Wave 5G and other communication systems. At the present the overflow of data has primarily been attributed to streaming video but the emergence of new unforeseen applications like holographic tv will introduce even more demand on bandwidth2. To support good user experience, the 5G network will also have to reach unique levels of flexibility and intelligence; spectrum regulation will need to be reviewed and enhanced, energy and cost efficiencies will become even more serious concerns4.The fifth generation (5G) technology standard for broadband cellular networks is needed to accommodate the exponential growth in the wireless data traffic driven by the proliferation of smart devices and live streaming. With advancement in technology, it is anticipated that the 5G wireless system will need to cope with a huge amount of data traffic resulting from high-resolution video streaming, tactile Internet, IoT based remote monitoring, real-time control applications and connected devices and vehicles. Billions of devices will be connected to the global IP network which will be a challenge and will require a lot of bandwidth5. The prominent frequency bands for the 5G mm-Wave based cellular communication are 24–28 GHz, 37–40 GHz, and 64–71 GHz7. This will inevitably raise new challenges10 including the free space path loss and hardware impairments. The amount of array gain required to compensate free space path loss becomes higher with the increase in frequency and the impact of atmospheric absorption due to carbon dioxide, oxygen and rain/fog/snow attenuation will impact significantly on service coverage compared to existing broadband mobile systems. Consequently, the number of antennas required needs to be increased. This increases the associated electronics with each antenna elements making the overall array very expensive compared to its lower frequency counterpart11.The current 5G wireless communication system is not equipped to meet this explosive growth in traffic demand hence the need for 5G systems to operate at millimeter-waves to acquire much larger bandwidth to accommodate the intense data traffic. The frequency bands below 20 GHz are highly congested, so a shift to higher part of the frequency spectrum is inevitable16. At this frequency band the atmospheric losses are effectively minimal17. This should to some extent ease the design of 5G mm-Wave systems to achieve specifications of high bandwidth and data rate requirements. Reported in5 is an antenna operating at 38 GHz, i.e., one of the 5G, which could handle the atmospheric attenuations challenge at mm-wave transmission. The peak gain of the antenna is 10 dBi. An array antenna comprising four elements is reported in12 which operates at the 38 GHz band. This antenna array is shown to have a gain of greater than 12 dBi, which is sufficient for the 5G mobile systems. Similarly, the antenna presented in13 operates over the 5G mm-Wave band, i.e., 37–40 GHz. The array configuration exhibits gain of up to 12 dBi. Antenna array in14 covering the 37–39 GHz band has a maximum gain of 8.81 dBi. The concept of metasurface would be helpful for the design of base station antennas operating over 5G mm-Wave spectrum17. Metasurface is used to enhance the antenna’s performances in18 but the maximum gain is limited to 8.91 dBi. The antenna presented in19 can cover the band from 37.1 to 38.1 GHz, but no gain enhancement is implemented to handle the atmosphere attenuations challenge. Likewise, an array structure is implemented in24 for the similar antenna element presented in18 to improve the gain. But still, maximum gain of nearly 12 dBi is achieved which is not sufficient for base station communication. Thus, the antenna element design presented in24 is considered in this work to develop such an array structure which could provide enough gain for the base station communication.A lot of work has been done recently in developing antennas for 5G wireless systems covering the frequency band 37–40 GHzThe antennas mentioned above are dedicated for the 5G mobile devices but cannot be adopted for base station applications because of their low gain. Consequently, in this work, we propose a novel antenna array suitable for 5G mm-wave base station applications. Each radiating structure in the antenna array consists of a 2 × 2 array. This configuration is used to boost the overall gain of the array at the mm-wave band. The gain of the proposed 8 × 32 antenna array is more than 20 dBi at 37.2 GHz, which is suitable for application in future 5G mm-Wave base stations.24. This antenna configuration was chosen for its impedance matching characteristic compared to the conventional rectangular patch antenna24. However, it has a smaller effective aperture size than a conventional rectangular patch antenna operating at the same center frequency and hence a lower gain. The antenna was designed to resonate at 37.2 GHz. The antenna is constructed on a Rogers 5880 dielectric substrate with a relative permittivity of 2.2, a thickness of 0.254 mm, and loss tangent of 0.0009. The size of the substrate used is 10 × 6 mm2. The dimensions of the antenna are listed in Table The geometry of the antenna used in the design the 5G mm-Wave arrays, i.e., 8 × 8, 8 × 16, and 8 × 32, is shown in Fig. 2.The proposed antenna was used to implement an 8 × 8 antenna array, shown in Fig. 2. The dimensions of the 8 × 8 array is given in Table The 8 × 8 array was extended to 8 × 16 array, which consisted of 32 antenna elements. This was achieved by using a power divider to split the input signal into two equal phase output signals that are applied to the two 8 × 8 array, as shown in Fig. 2. The dimensions of the F-parameters annotated in Fig. The 8 × 16 array was extended to 8 × 32 array using the above principles, as shown in Fig. 2 however this increases to 30 × 110 mm2 for a 8 × 32 array. The 8 × 32 operates at the 5G mm-Wave band. The fabricated prototype of the 8 × 32 array is shown in Fig. The simulated reflection coefficient of the single element antenna and the three antenna arrays, i.e., 8 × 8, 8 × 16 and 8 × 32, are shown in Fig. The radiation patterns of the proposed antenna and the three arrays of different matrix size using the antenn are compared in Fig. 21 a dual band 5G multiple input multiple output (MIMO) antenna is reported that operates at 28 GHz and 38 GHz. The sub-array in21 consists of four gap-coupled microstrip slotted dual band patch antennas with corporate microstrip feed network. This sub-array exhibits a realized gain of 12 dBi at 28 GHz and 13 dB at 38 GHz. The antenna array in22 is based on stacked square ring patch arrangement. This antenna has a sharp roll-off and a filter like response between the operating bands due to the strongly coupled resonators. The antenna operates from 24.25 to 29.5 GHz and 37 to 40 GHz. This antenna array has a realized gain of between 5 and 6 dBi. In23 the authors have created an antenna array by stacking multiple split-ring resonators that interact with each other via aperture coupling. This antenna array operates from 24.25 to 29.5 GHz and has a realized gain between 5 to 6 dBi.Over recent years numerous mm-wave antenna array designs have been investigated. Table 24 consists of a two-antenna array combination where each antenna array consists of four elements which are arranged orthogonally with respect to each other. The antenna array has a gain of 12.8 dBi at 37 GHz. The 29 GHz magneto electric (ME) dipole array in26 is realized by the method of connecting and cutting patches. In27 a printed ridge gap waveguide 4 × 4 Butler matrix is shown to operate over the frequency range from 26.9 to 33.3 GHz with gain variation from 10.2 to 11.35 dBi. The array element in the broadband end-fire antenna in28 consists of a horizontally oriented printed electric dipole and a vertically aligned tapered slot radiator, where the two orthogonal radiated electric-field components are excited simultaneously. A parasitic director is introduced near the printed dipole to compensate for the gain degradation of the dipole element at higher frequencies.The MIMO antenna array in29 the 8 × 8 antenna array consists of a block of metallic array antenna sitting directly on the top surface of a printed circuit board (pcb) with electrical connections between them. The metallic array antenna is a metallic structure with 64 units of antenna radiating elements surrounded by a ring of metallic wall. The pcb provides the feeding mechanism for all 64 units of antenna radiating elements on the metallic array antenna. This antenna has a gain of 23.95 dBi at 26 GHz. The 8 × 8 element antenna array reported in30 uses a stacked multilayered pcb. It uses 16 commercial quad-core transmitter and receiver integrated circuits to independently control the phase and amplitude at each radiating element. The array designed at 39 GHz frequency band has a gain of 22 dBi. Reported in31 is an 8 × 8 aperture-coupled microstrip patch antenna array. It constructed from two substrate layers that are separated with an air gap. The patch antenna on the top layer is excited through a ground slot implemented on the top of the bottom substrate layer with a feedline on the bottom layer. This array is designed to operate over the 22–27 GHz frequency band with 22 dBi gain.In32. The 8 × 32 antenna array in32 consists of a square patch antenna designed to operate at 35 GHz. The antenna array is reported to have a gain of 25.7 dBi. Although the gain in32 is higher than the proposed 8 × 32 antenna array however it has a much larger size by a factor of 3.6.The work reported on 8 × 32 patch antenna arrays is scarce however the authors managed to find one paperResults presented in the paper show the feasibility of a novel patch antenna configuration in the design of high-gain antenna arrays for application in future 5G mm-Wave base stations. The gain provided by the proposed singular antenna is 7.6 dBi. Although, this gain is suitable for mobile communication devices but it is not sufficient to overcome path loss and atmospheric loss experienced my millimeter wave signals at the 5G base station. Hence, a 64-element array antenna was implemented using the proposed antenna. This antenna array is shown to have a bore side gain of 21.2 dBi at 37.2 GHz, an angular width of 4.1 degrees and side lobe levels less than − 10 dB. This antenna has a significantly smaller form factor than 8 × 32 antenna array reported to date.All of the figures, materials, and data within the manuscript are original and owned by authors.


As the Internet of Things (IOT) becomes more widely used in our everyday lives, an increasing number of wireless communication devices are required, meaning that an increasing number of signals are transmitted and received through antennas. Thus, the performance of antennas plays an important role in IOT applications, and increasing the efficiency of antenna design has become a crucial topic. Antenna designers have often optimized antennas by using an EM simulation tool. Although this method is feasible, a great deal of time is often spent on designing the antenna. To improve the efficiency of antenna optimization, this paper proposes a design of experiments (DOE) method for antenna optimization. The antenna length and area in each direction were the experimental parameters, and the response variables were antenna gain and return loss. Response surface methodology was used to obtain optimal parameters for the layout of the antenna. Finally, we utilized antenna simulation software to verify the optimal parameters for antenna optimization, showing how the DOE method can increase the efficiency of antenna optimization. The antenna optimized by DOE was implemented, and its measured results show that the antenna gain and return loss were 2.65 dBi and 11.2 dB, respectively. In IOT applications, many wireless communication modules are used. The antenna is a wireless communication module and an essential circuit component. Antenna characteristics are directly related to circuit applications and application scenarios, and therefore, antennas often must be customized. Consequently, antenna design efficiency has a direct relationship with the time to market of a product and thus is highly influential in product development.To increase the efficiency of antenna design, extensive research has been conducted on antenna synthesis and design optimization. GA and the method of moments were used to design a broadband patch antenna, increasing the antenna by 20% . In anotChen and Ku  stated tIn this study, a design of experiments (DOE) method is proposed to improve the efficiency of antenna optimization. Some layout parameters of the patch antenna are analyzed with RSM to optimize antenna gain and input return loss performances. The results show that the estimation of antenna performance with DOE to optimize an antenna is more efficient than optimization with the EM simulation tool. Performance of the antenna optimized by DOE was evaluated, and the measured results show that the antenna gain and return loss were 2.65 dBi and 11.2 dB, respectively.This research focused on the gain of a patch antenna. The aim was to shorten the time spent developing the antenna. We used the experimental design method to identify crucial factors and determine the scope of their influence, and thus find the optimal factor configuration and gain value so that the reflection loss is less than −10 dB. We then used antenna simulation software to verify the results. Traditionally, antenna design has involved using computer-aided engineering simulation software and the finite element method to simulate the electromagnetic field, and the characteristics of the radiation field of the designed antenna are then obtained. However, this method is excessively time-consuming, reducing the efficiency of antenna design. To shorten the development time and increase efficiency, this study (i) used a systematic experimental design method to determine the important factors and the scope of their influence, in addition to performing factor programming experiments, (ii) used RSM to determine the optimal factor combination, and (iii) employed antenna simulation software to verify that the identified combination of factors is optimal.In an antenna, radio waves begin at the internal feed. They then pass through the conductor between the chip and ground excitation from the radio frequency electromagnetic field, and then proceed through the patch around the ground surface and the gap to the outside. The size of the conductor and ground surface can be adjusted to achieve the target resonant frequency band.Rectangular metal radiation surface: located in the top layer in the middle of the rectangle, it sends and receives signals.Insulating substrate: located in the middle layer, it consists of insulation material with a dielectric constant of 4.4 .Ground metal surface: located in the bottom layer; for copper, the area must be larger than the rectangular metal surface.Coaxial probe: a small round coaxial conductor, where the metal surface is connected to the ground and the internal surface is connected to the metal surface. This feed is within the volume of the feed surface, and the energy through the feed surface flows into the antenna.Shorting pins: usually connected to two media, they may gain but contribute to an increase in bandwidth.A microstrip antenna is situated in a thin substrate with a thin layer of metal as a ground surface. Photolithography corrosion can be used to shape the metal patch, which acts as a radiating surface, and then a microstrip line with a coaxial probe is used for the patch feed. Microstrip antennas can be circular, rectangular, or ring-shaped. This study used a rectangular microstrip antenna, with the following characteristics :RectanguTwo antenna design parameters we used as the design parameters for optimization of antenna gain and reflection loss.Antenna gain can be used to measure the antenna’s directivity, which refers to the capacity of the antenna to transmit signals in a specified direction. Usually, the greater the effective area of the antenna is, the higher is the gain. Gain is measured in dBi units.Reflection loss concerns the feedback component of the feedback signal. The antenna reflection loss should be as small as possible, and at least −10 db. The smaller the reflection loss is, the greater is the signal input, and thus, the greater the radiation power.The Advanced Design System  was used for antenna design simulation . KeysighMinitab 18 statistical software was used to execute the experimental design using RSM. RSM supplements the prediction model by combining mathematical and statistical methods so that the experimental area can be adjusted according to the experimental scope to determine the ideal response value.First, we determine the variables that are influential to the system, and then we select an appropriate level as the initial value of the experiment. A two-level factorial experiment is performed initially. Many parameters often must be considered in the design of an antenna, and the responses are usually affected by the main effect and low-order interaction. A two-factor interaction is therefore appropriate for the model . The forWhere If the first-order regression model is appropriate, the steepest ascent method is used to search an area where the optimal response value should be located. The purpose of this method is to search for the best point.At this point, the curvature of the real response surface increases, which means that the first-order model of the best solution can no longer be applied and the second-order model must be used. In general, the central composite design (CCD) is employed as the second-order model because it is very efficient.For the CCD, a k-factor experiment with a  example . After aAfter accepting the second-order regression model, we can begin the analysis using RSM. The second-order function can be represented by the matrix shown in Formula (3) to identify the stationary point. The stationary point may be the maximum, minimum, or saddle point. According to the extreme theorem, if the solution of function (4) at the extreme value of the differential is 0, we can obtain Formula (5). If the characteristic root is negative, the stationary point has the maximum value; if it is positive, the stationary point has the minimum value; otherwise, the stationary point is the saddle point:iy is converted to the desirability function di, where 0 ≤ id ≤ 1. When the response value iy reaches the target, di is equal to 1. If it exceeds the acceptable area, then id is equal to 0. The overall desirability function is D = /m1, where m is the number of responses, iy.RSM often involves multiple response values; therefore, the desirability function can be used to obtain multiple response optimizations. The desirability function can identify the best objective function of a single response value through mathematical transformation based on the upper and lower bounds of the response variable and its target value . Each reThe next section describes the application of the antenna design. Factor experiments were performed to conduct the initial fitting, and RSM was employed to identify the stationary point. Finally, the response value was optimized to find the optimal combination of parameters.The shape of the antenna is shown in 6-1) were performed.The first experimental measurement was based on the current size of the antenna, and we defined the maximum range within which the factors can increase or decrease as the upper and lower limits of two levels. These are shown in Minitab 18 statistical software was used to conduct the regression analysis, in conjunction with backward elimination. First, all antenna design parameters were entered into the regression model. The smallest explanatory antenna design parameters were then excluded until all significant changes were removed.In the preliminary experiments on the gain model, factor A was deleted from the model, indicating that this was not a significant factor in terms of gain. Additionally, we tested the RL model. No factor was deleted; therefore, every factor was significant in this regression model.To find the optimal combination of antenna design parameters, main-effect graphs were drawn. The main effects of the six factors on gain were identified. Factor A was not significant; factor B, D, E, and F were the smaller the better, and factor C was the larger the better. Regarding the main effects on RL, factors B, D, and E were the larger the better, whereas factors C and F exhibited no significant effects.The preceding analysis indicates that factors B, D, and E exhibited opposite trends with regard to gain and RL responses. Factor C was the larger the better for gain, but for RL, it exhibited no significant trend. Conversely, factor F was the smaller the better for gain, but for RL, it exhibited no significant trend. These results were employed as benchmarks that enabled us to adjust the factorial range in subsequent experiments.Because the preliminary experiment was not ideal, we adjusted several factors and their ranges for the second trial, as shown in We used background elimination to conduct the regression analysis. We found that all factors were significant in the model for gain. The R-sq was 66%, indicating that the model was appropriate. The formula is presented as follows:All factors were significant in the model of RL. The R-sq was 93.21%, indicating that the model was appropriate. The formula is presented as follows:The objective of this study was to maximize gain and reduce RL to less than −10 dB. Therefore, in the response optimizer in MINITAB, the target of gain was set as the maximum, and the target of RL was set as the minimum. The target was to achieve maximum gain with an RL less than −10. Using Minitab 18 statistical software, we found that the overall desirability function of the prediction model was 0.9271, the desirability function for gain was 0.87864, and that for RL was 0.97823. According to the model, the optimal response values are a gain of 10.6203 and an RL of −27.0495. In these conditions, factor A is 4.5, factor B is −1.5, factor D is −0.4, factor E is −0.1, factor C is fixed at 0.9, and factor F is fixed at −2.4.Minitab18 was then used to generate a set of feasible solutions in order to determine the values for practical needs. Antenna simulation software was used to verify this set of data, and the results in Using the ADS electromagnetic field simulation software for analysis, the antenna reflection loss and gain characteristics were obtained. As shown by the dotted lines in Based on the optimized design of the antenna structure shown in The AMS-8600 antenna measurement system was used to measure the gain performance of the antenna, The findings of this study show how the experimental design method works to increase the effectiveness of antenna design. Using Minitab 18 statistical software, the ideal characteristics of the antenna were identified by setting the target of gain as the maximum and the target of reflection loss as the minimum. The results were then verified using software for antenna simulation, and the fabrication and measurement of the antenna provided additional confirmation. The simulation results and the measured reflection coefficient and gain agreed very closely, proving the accuracy of the optimization technique. Even though parasitic effects caused the measured gain to deviate slightly from the simulation, the results still offer important information about the antenna’s functionality. Overall, the method described in this study can be used to design different kinds of antennas, offering a useful and effective way to maximize antenna performance.In this paper, we propose a method for improving the efficiency of antenna design. Using the experimental design method, which involved setting the size of the antenna and the antenna design parameters, we quickly determined the optimal characteristics of the antenna, and then demonstrated the feasibility of this method using simulation software. In this research, an E-type plate antenna was used as an example, and the antenna was designed at 5.8 GHz. The measured reflection coefficient was 11.2 dB and the measured gain was 2.65 dBi. This method can therefore be applied to antenna design to improve efficiency.