This article presents an ultra-wideband (UWB) monopole antenna with triple band notch characteristics. The proposed antenna consists of an octagonal patch, fed with a 50 Ω line, which occupies a compact size of 40 mm × 29 mm  and resonances at a relatively low frequency (2.94 GHz). Specifically, an L-shaped stub, an inverted C-shaped slot, and a pair of U-shaped resonating structures are introduced into the design, which allow antenna to generate three band notches at 3.22–3.83 GHz, 4.49–5.05 GHz and 7.49–8.02 GHz, corresponding to WiMAX band, Indian national satellite (INSAT) band, and X-band satellite frequencies, respectively. In the center of the notched band, the antenna has lower efficiency and gain, essentially indicating that the antenna has good interference rejection performance. To evaluate its performance, the proposed antenna has been fabricated and measured, and the relevant functional parameters, such as S-parameters, voltage standing wave ratio (VSWR) and radiation property, have been studied. Ultra-wideband (UWB) technology has great potential for application in wireless communication because of its extremely low transmission power and high data rate . In receOver the years, researchers have proposed various methods to design the band-notched UWB antennas ,10,11,12The following are the novel discoveries and contributions of this work:1. The controllable triple-notch frequencies are achieved at the WiMAX, INSAT and X-band satellite frequency bands.2. The proposed antenna integrates multiple forms of notch structures with different shapes and techniques.In this work, a planar UWB monopole antenna is designed, fabricated and tested. The dimensions of the stub and slot are varied to achieve the desired stopband center frequency. Furthermore, three different band-notched designs are provided for illustration, along with a description of the design concept. Lastly, the details of antenna measurement and simulation results are presented, which demonstrate a successful band-rejection capability for all three proposed band-notched designs. Above all, the proposed antenna can be a potential option for specific devices operating in WiFi 6E band.r = 4.4) with a thickness of 1.6 mm. Moreover, a 50 Ω microstrip-line was fed by an SMA connector. The geometry and configuration of the proposed antenna have been designed and optimized using HFSS 18.0, as shown in The proposed antenna is fabricated on a common and low-cost FR4 substrate Qi=fiThe details of all the stages of the design evolution process are given below:Step-1 includes an octagonal radiation element with a rectangular ground structure a. This af + Lp + L1 − Ws. Here, L-shaped stub acts as an open-circuit transmission line that shorts the antenna at the relevant frequency. Thus, the effective transmission path of current is changed [Step-2 introduces an L-shaped stub at the upper left corner of the octagonal patch b. The di changed , causingBased on the design of step-2, an inverted C-shaped slot is etched onto the center of the octagonal patch, which defines step-3 c. At cerThe specifics of the rejection bands for the design of each stage have been listed in c is the speed of light, f is the notched center frequency, effε is the effective dielectric constant.An approximate size of the notch structure can be assumed as :(4)L=c2fNext, a transverse U-shaped resonator is placed symmetrically on both sides of the feed line to shield the antenna from X-band system interference, which is step-4 d. Here, s, the corresponding notched band becomes smaller, while the operational band remains unchanged. Similarly, by adjusting the size of inverted C-shaped slot opening, the entire second notch band can be shifted from a low frequency to a high frequency. Moreover, as the L3 increases, the range of the notch band becomes wider and the center frequency of the notch band shifts down. The dimension of Lf effects the phase of the antenna, while L1 and W1 determine the area of the radiation patch. When they increase or decrease, the resonant frequency shifts significantly in the range of 5.2−7.4 GHz and 8.1−11.0 GHz. Accordingly, it can be concluded that, with the proposed design approach, the notched frequency bands can be easily achieved and controlled to meet the practical requirements by merely adjusting the dimensions and locations of the resonating elements. Meanwhile, it is worthwhile noting that changing the parameters of resonators (notch elements) affects only the notch bands, and the return loss in the rest of the UWB frequency band remains virtually unchanged.Next, the effects of different geometrical parameters of the proposed antenna on the band notch characteristics are studied, as portrayed in To validate the proposed antenna design, an ultra-wideband antenna was successfully fabricated and tested according to the dimensions listed in Furthermore, radiation patterns are plotted in In this work, a compact planar monopole ultra-wideband antenna with anti-interference characteristics has been presented for UWB applications. Interestingly, the proposed antenna provides a wide impedance bandwidth, ranging from 2.70 GHz to 11.06 GHz. In addition, there was a discussion of three rejection bands around 3.22~3.83, 4.49~5.05 and 7.18~7.84 GHz for the applications of the WiMAX, INSAT and X-band. These were created by introducing an L-shaped stub in the radiation patch, a pair of U-shaped parasitic elements beside the feed line, and an etched inverted C-shaped slot. Additionally, the designed antenna had a simple structure and easy fabrication process. The antenna also possessed an acceptable peak gain and efficiency, demonstrating that the proposed antenna was certainly applicable in miniaturized devices for the the operation of UWB communication systems.


S11 ≤ −10 dB was achieved by regulating the radiator and the partial ground that was placed on the second side of the antenna substrate. The CP was achieved when excited two modes with the same amplitude and a 90° phase difference. This could be generated by regulating the slots’ dimensions in the ground plane. Moreover, a quadrilateral-shaped parasitic strip placed on the second side with the partial ground was utilized to extend the 3 dB axial ratio (AR) bandwidth. The suggested structure is simulated, prototyped, and measured to confirm the desired requirements with a total size of 30 × 32 mm2 (0.4 × 0.42 λ0 at 4 GHz). The tested outcomes have a bandwidth of S11 ≤ −10 dB (81.25%)  and a 3 dB axial ratio (AR) bandwidth (30.7%) . The antenna’s different parameters are discussed, which recommend the suggested antenna to be used in UWB, sub 6 GHz, and WLAN wireless applications.A circularly polarized (CP) and wide-band monopole antenna with a miniaturized size is suggested in this study. The suggested structure is composed of a U-shaped radiator on the front side, a partial ground plane with two rectangle slots, and a quadrilateral-shaped parasitic strip on the back side of the FR4 substrate. A wide-band operation with  Recently, wireless communication systems of a small and compact size that operate in a wide-band operation need compact and wide-band antennas that can be easily integrated with them ,2. CP anIn modern wireless systems, for example, wide-band circular polarization (CP) antennas, radio frequency identification (RFID), and global positioning systems (GPSs) are recommended and utilized. Recently, wireless systems have been operated at different frequency bands. Therefore, wide-band CP antennas are considered a good choice to reduce the system’s complexity and price ,7,8. MicMicrostrip antennas have several advantages, such as low price, simple design, small size, ease of integration, and simplicity in CP realization. Therefore, the microstrip antenna is used in several communication systems. However, conventional CP antennas are produced with a small 3 dB axial ratio with a large size, which cannot be suitable for wide-band systems ,10. For 2 and peak gain of 3.6 dB. Additionally, the ground plane is modified to increase the 3 dB AR bandwidth to 53.92% from 4.28 to 7.44 GHz. The antenna has a simple structure with a wide-band 3 dB AR bandwidth and is utilized for WLAN communication.In , a paras2 and a peak gain of 11.2 dB. Additionally, the antenna has a 3 dB AR bandwidth from 1.5 to 7.5 GHz (133.33%). While the AR in [The UWB CP operation generated using a sequential phase feed is investigated in . It is c2 and a peak gain of 6.55 dB. Additionally, the ground plane is modified to increase the 3 dB AR bandwidth to 104.7% from 2.05 to 6.55 GHz. The antenna has a simple structure with a wide-band 3 dB AR bandwidth and is utilized for WLAN communication; however, it has a large size.In , a broad2 and a peak gain of 2.2 dB. The antenna has a simple structure with a narrow 3 dB AR from 1.572 to 1.578 GHz (0.4%). The antenna has a narrow band of 3 dB AR bandwidth with a large size. A 3D with complex structure microstrip antenna is introduced in [2 and a peak gain of 4.3 dB. The antenna has a 3 dB AR bandwidth of 3%. The antenna has a complex structure with a large size.A microstrip antenna utilizing a fractal-defected ground structure is discussed in . The antduced in . An L-sh2. The antenna has a peak gain of 7.2 dB. Additionally, the antenna has a 3 dB AR bandwidth of 27.5% from 4.55 to 6 GHz. The antenna has a complex structure and it is utilized for WLAN communication. A slot antenna with a CPW feed is introduced in [2 and a peak gain of 3.8 dB. Additionally, the antenna has a 3 dB AR bandwidth of 58.6% from 2.85 to 5.21 GHz. The antenna has a simple structure with a large size and it is utilized for WLAN communication.In , a microduced in . The ant2 and a peak gain of 4.8 dB. Additionally, it has a 3 dB AR bandwidth of 4% from 4.25 to 5.95 GHz. The antenna has a simple structure with a large size and works for wireless communication. Parasitic strips are investigated in [2 and a peak gain of 2.9 dB. Additionally, it has a 3 dB AR bandwidth of 4% from 2.25 to 2.35 GHz. The antenna has a simple structure with a large size and works for satellite communication.Moreover, some modifications are utilized in the monopole antenna to increase the 3 dB AR bandwidth. Lumped capacitors are used in . An invegated in ,18. A Y-gated in . The ant2 and a peak gain of 2.45 dB. Additionally, it has a 3 dB AR bandwidth of 32% from 1.42 to 1.97 GHz. The antenna has a simple structure with a large size and works for satellite communication. A parasitic open-loop resonator is added to a rectangular monopole antenna to generate the CP operation. The antenna is operated from 1.48 to 4.24 GHz (96.5%). The 1 mm FR4 substrate is utilized in the design. The antenna has a size of 50 × 55 mm2 and a peak gain of 3.5 dB. Additionally, the ground plane is modified to increase the 3 dB AR bandwidth to 63.3% from 2.05 to 3.95 GHz. The antenna has a simple structure with a wideband 3 dB AR bandwidth and a large size.A slot antenna with an L-shape is suggested in . An L-sh11 ≤ −10 dB (81.25%)  and a 3 dB axial ratio (AR) bandwidth (30.7%) . The designed antenna keeps the same features as a common turnstile antenna, such as simplicity, low cost, compact, low profile, and broadband CP antenna. All simulated results are extracted utilizing CST software. The suggested antenna can be used in UWB, sub 6 GHz, and WLAN wireless applications.A wide-band CP antenna is introduced in this work. A U-shaped radiator on the front side, a partial ground plane with two rectangle slots, and a quadrilateral-shaped parasitic strip on the back side are utilized to produce the CP feature. The CP is achieved when we excited two modes with the same amplitude and a 90° phase difference. This can be generated by regulating the slot’s dimensions and adding a quadrilateral-shaped parasitic strip to the ground plane. The tested outcomes have a bandwidth of SThe paper is organized as follows: (I) The literature review is introduced. (II) The configuration of the antenna is investigated. (III) The simulation and measurement outcomes are provided. (IV) The conclusion is presented.3 and εr = 4.4, tan δ = 0.025 was used as a substrate. The suggested antenna was connected to a 50 Ω microstrip feed line with an optimal position to enhance the matching of the antenna. An L-shaped connected to a C-shaped mirror to compose a suggested U-shaped radiator was added to the front side. Moreover, a partial ground plane with two rectangle slots and a quadrilateral-shaped parasitic strip was added to the back side of the substrate. The two rectangle slots were cut from the edge of the ground plane to achieve CP modes at high frequencies. The wide CP generation was achieved by adding a quadrilateral strip to the bottom of the substrate, as illustrated in The proposed wide-band CP monopole antenna with the complete geometrical configuration is shown in The suggested structure was passed through four stages to achieve the desired final design. The four-step antenna design procedure is illustrated in The CP radiation can be obtained by achieving two orthogonal modes with equal amplitudes and a 90° phase difference. However, the conventional monopole antenna, as shown in Finally, in Ant. #4, the back side of the structure was added with a quadrilateral shape, as shown in The CP generation behavior could be understood by displaying the antenna distribution current at different orthogonal phases at 5.8 GHz, as presented in L4 on the antenna’s performance. By increasing the length of L4 from 1.45 to 7.45 mm, the antenna was operated from 4 to 9 GHz with a good impedance bandwidth as shown in L4 = 1.45 mm, the 3 dB AR bandwidth was extended from 4.48 to 4.8 GHz. Additionally, when L4 = 4.45 mm, the 3 dB AR bandwidth was extended from 4.48 to 6.11 GHz. Finally; when L4 = 7.45 mm, the 3 dB AR bandwidth was extended from 4.7 to 5.5 GHz. The length of L4 was chosen to be 4.45 mm.From the previous section, it can be noticed that the dimension of the quadrilateral-shaped strip can affect the 3 dB AR. Therefore, a parametric study was utilized to show its effect. L4 = 4.45, the antenna was operated from 4 to 9 GHz with a good impedance bandwidth as illustrated in W6 = 2 mm, the 3 dB AR bandwidth was extended from 4.48 to 6 GHz. Moreover, when W6 = 2.5 mm, the 3 dB AR bandwidth was extended from 4.48 to 6.11 GHz. Finally, when W6 = 3 mm, the 3 dB AR bandwidth was extended from 4.48 to 5 GHz and from 5.6 to 6.2 GHz. The length of W6 was chosen to be 2.5 mm. Finally, by elaborating on the parametric study outcomes, the final dimensions achieved the desired bandwidth from 4 to 9 GHz and satisfied the 3 dB AR bandwidth from 4.48 to 6.11 GHz.Furthermore, 3, and εr = 4.4, tan δ = 0.025 were used in the fabrication. Additionally, it was tested using a vector network analyzer (R&S ZVA 67) to show the reflection coefficient S11, as shown in S11 ≤ −10 dB, while the simulated outcomes illustrate that the antenna is operated from 4 to 9 GHz. The two results display good matching between them, with some slight deviations between them due to the fabrication tolerance and SMA soldering process.The photolithography method was utilized in the fabrication process. xz (φ = 0°) and yz (φ = 90°) planes. The co- and cross-polarization results at 5.5 and 5.8 GHz are shown in The setup of the far-field results was conducted inside an anechoic chamber, as illustrated in The gain of the antenna was measured, as illustrated in Finally, the suggested design was compared with others to evaluate the novelty of the work, as presented in 0 at 4 GHz).A miniaturized size and wide-band CP monopole antenna was suggested, fabricated, and tested. The suggested antenna was 30 mm × 32 mm  and an AR bandwidth of 30.7% . Based on the achieved outcomes, it can be suggested that the antenna is considered a good choice for several wireless systems, such as UWB, sub 6 GHz, and WLAN applications.A U-shaped radiator on the front side, a partial ground plane with two rectangle slots, and a quadrilateral-shaped parasitic strip on the back side were utilized to produce the CP feature. The CP was achieved when we excited two modes with the same amplitude and a 90° phase difference. This could be generated by regulating the slot’s dimensions and adding a quadrilateral-shaped parasitic strip to the ground plane. The antenna’s different parameters were discussed and investigated. The tested outcomes had a bandwidth of


This paper presents novel approaches for reducing the mass of the classical short backfire (SBF) antenna by using additive manufacturing and structural perforations. We first investigated techniques to create a 3D-printed structure with a conductive coating material. This approach resulted in a significant mass reduction (70%) compared with the conventional metallic structure. We performed parametric simulation studies to investigate the effects of the manufacturing process and showed that there was practically no difference in the performance. The largest source of error was the surface roughness and the conductivity of the metal paint. In a second design, we created perforations in the structure to further reduce the mass. We performed parametric studies to optimize mass reduction and to characterize the effects of the perforations and the surface roughness introduced during the 3D-printing process on the antenna. Antenna prototypes were fabricated and tested. The masses of the perforated 3D printed antenna were approximately 30% and 20% of the original aluminum design, respectively . The good agreement among the original design, simulation, and measurements demonstrated the effectiveness of the approach. In caseeflector . Drones Traditional SBF, reflector, and horn antennas can range from 500 g  to 8 kg The first approach to reducing the SBF antenna’s mass involved fabricating the antenna using additive manufacturing techniques, also known as 3D printing. Over the last few years, antenna and electromagnetic constructions have been printed using various 3D-printing methods, including photonic polymerization, stereo lithography, micro stereo lithography, Polyjet printing, and fused deposition modeling (FDM). Each technique provides advantages and disadvantages that are best suited to producing various types of antennas. For instance, the FDM technique, which extrudes filament materials from a heated nozzle, is typically the most affordable 3D-printing process while also offering the advantages of a large selection of filaments and the ability to print with many materials at once. Laser or UV light is used in the stereolithography (SLA) and polymerization technique to cure resin components into the desired 3D shape. Therefore, they can attain a resolution significantly higher than that of FDM, making them suitable for producing smaller structures. Because the materials used in these printers are essentially a light-curable resin, they cannot be altered to the same degree as FDM filaments .For most metallic antennas, such as the SBF antenna, a conductive surface is usually required to interface with the input port. Poor conductors, which are suitable for 3D printed electrical circuits, are not appropriate for antenna applications since the losses negatively influence radiation efficiency. To fabricate metallic antennas, specialist metal 3D printers that can directly print metallic materials such as aluminum, steel, titanium, and cobalt through direct metal laser sintering (DMLS) provide a good option ,10. An aThe perforation technique is another method used for the weight reduction of reflector and horn antennas. This technique has been achieved in the literature by either creating holes/slots in the antenna ,14,15,16The work presented in this paper aims to reduce the weight of SBF antennas while meeting the operation specifications of a C-band remote sensing system. This is done by implementing an improved aluminum SBF antenna prototype using the 3D-printing technique in the first candidate design. The perforation technique was applied to the second antenna to further reduce the overall weight. A simulation study was completed to characterize the performance, and a prototype was fabricated using a 3D printer to verify the concepts.The proposed SBF antenna designs are based on the improved SBF antenna presented in  and show0 = 54.54 mm in free space. The antenna is expected to have a minimum impedance bandwidth of 500 MHz and a minimum gain of 14 dBi within the frequency band of operation to perform efficiently. In this study, the antennas presented were designed to weigh less than 50% of the aluminum prototype.Based on previous research on conventional reflector antennas, we looked for ways to reduce the mass of the SBF antenna by using lighter materials, such as thermoplastics, for the design process. Designing with thermoplastic cannot be easily done when using conventional milling fabrication and often requires a 3D-printing fabrication process. Polylactic acid (PLA) and acrylonitrile butadiene styrene (ABS) were used as the 3D printer filaments for building the antenna prototypes presented in this manuscript because of their low cost and strong mechanical characteristics. The metal coating was applied using the aerosol spray technique with the MG Chemicals Super Shield nickel conductive coating and the MG Chemicals Super Shield Silver Conductive Paint. These methods and materials were chosen because of their cost-effective and fast application process. They have also been reported to produce excellent results .Because 3D printing is a layer-by-layer process, surface roughness is introduced and can be detrimental to the performance of high-frequency antennas. Consequently, it is crucial to simulate the effects of different surface roughness values on the performance of the SBF antenna. The body of the antenna was modeled in HFSS using a material with a relative permittivity of 2.8 and a loss tangent of 0.005 to roughly approximate the properties of the ABS plastic for the simulation of the 3D-printed SBF antenna. . To simu11 results of the aluminum and the 3D-printed prototype for SR values of 0.6 Ω/sq, 3 Ω/sq, and 10 Ω/sq. From this figure, we can see that the impedance bandwidth of the SBF antenna increased with the surface resistance. The 10 dB impedance bandwidth of the aluminum was calculated to be 1.8 GHz (37.5%), while those of the 3D-printed antenna were 1.84 GHz (38.5%), 1.86 GHz (39%), and 1.92 GHz (40%) for SR values of 0.6 Ω/sq, 3 Ω/sq, and 10 Ω/sq, respectively. This increase in impedance bandwidth with increasing roughness is supported by the equations given in [Surface impedance increases with surface roughness while decreasing with the metal layer’s conductivity and thickness . Given tgiven in , showingThe peak realized gain results were calculated for several surface impedance values. The summary of the results is given in The simulated worst-case cross-polarization ratio curves for the aluminum and 3D-printed antenna designs are given in 11); on the other hand, the gain decreased as the surface roughness and resistance increased. It was, therefore, important to keep it as low as possible by polishing the antenna before metallization and using a highly conductive metal spray. Comparing the weight of the ABS plastic (1.0 g/cm3) to that of the aluminum metal (2.7 g/cm3), the greatest advantage of the 3D-printing fabrication process compared to the conventional method is a 70% reduction in mass.In this section, the effects of surface roughness and resistance on the performance of the 3D-printed SBF antenna are studied and compared with the performance of the aluminum SBF antenna. We noticed that the increased surface roughness and resistance introduced by the 3D-printing and metallization processes did not have a significant effect on the reflection coefficient (S0 = 54.54 mm) in order to achieve weight reduction. Because the diameter of the holes is less than λ0, the transmitted EM wave does not radiate from these holes; hence, they do not significantly affect the performance of the antenna. Numerical analysis of the unperforated SBF antenna was implemented using HFSS (ANSYS Electronics Desktop 2021 R2©) in the previous section. An interesting phenomenon observed during the analysis was that most of the current density was concentrated in the main reflector, while the rim had a much lower current density. By taking advantage of this current distribution, portions of the rim were removed without disturbing the performance.The perforation technique is a relatively simple concept that involves purposefully creating a series of sub-wavelength slots in the antenna, meaning that the slots are significantly smaller than the antenna’s wavelength  were simulated to investigate their effects on the performance of the antenna, such as those on the gain, impedance bandwidth, and cross-polarization ratio.A plot of the peak gain and the impedance bandwidth as a function of the radius of perforation at an operating frequency of 5.5 GHz is given in 11 result for the perforated antenna (radius = 1 mm to 5 mm) is compared with that of the unperforated antenna in The SSimulation studies on the perforated SBF antenna showed that increasing the size of the perforations minimally decreased the impedance bandwidth. This was because of the surface current redistribution and impedance change around the waveguide aperture caused by the discontinuities introduced by the perforations. The peak gain at 5.5 GHz stayed almost the same for increasing perforation size, which is what was expected for perforation sizes that were significantly smaller than one wavelength. The gain of the antenna did not significantly change because the current distributions at the upper rim and aperture were not substantially affected. Consequently, the radiation pattern was not drastically changed either. As a result of the higher current density associated with this design, it was important to reduce the surface resistance caused by the 3D-printing and metallization processes to avoid extreme gain loss. Compared to the unperforated prototype, the removal of portions of the antenna’s rim in addition to the use of lightweight plastics for fabrication resulted in an 80% overall reduction in the antenna’s mass.The simulated E-plane and H-plane radiation patterns (θ = −180–180°) for the unperforated, nickel-coated, and silver-coated perforated SBF antennas are given in The Airwolf Axiom 3D printer was used to fabricate the 3D-printed prototypes at the University of Manitoba’s machine shop. The 3D-printing technique used fused deposition modeling (FDM). To make sure that the antenna was operating at optimum performance, the first post-processing step involved polishing the surface of the 3D-printed antenna by sanding it. Since the plastic was not conductive, the last step involved metallizing the model using a conductive metal spray paint. The entire antenna surface was covered by several layers of paint to achieve a good conductive and reflective surface.3) was used as the thermoplastic filament. After the antenna was printed, it was sprayed by hand using the MG Chemicals Super Shield Nickel Conductive Coating. The thickness of the metal coating was measured to be approximately 0.3 mm, which was much thicker than the skin depth of nickel at 5.5 GHz (0.07246 μm). A piece of foam material (30 mm thick) with an approximate dielectric constant of 1.05 was placed inside the main reflector cavity to position the sub-reflector at the required height. The fabricated solid 3D-printed antenna is shown in 11) of the 3D-printed antennas were measured using a Keysight PNA Network Analyzer (N5224B), and the measured far-field results were obtained in the Compact Antenna Test Range of the University of Manitoba. The S11 performance measurement of the waveguide adaptor (WR-159) was found to have complex impedances for frequencies below 5.2 GHz, which resulted in multiple oscillations that resulted in characteristic differences between the simulation and measurements , two copper tape strips were adhered to the aperture of the waveguide. This improved the impedance match by creating an inductive iris, and this approach was also applied to the 3D-printed antenna to obtain a similar result. The measured S11 plots for the aluminum SBF antenna and the simulated and measured S11 results for the solid 3D-printed prototype are provided in 11 results of the aluminum and the solid 3D-printed SBF antennas; the measured impedance bandwidths were 1.35 GHz (27.3%) and 1.38 GHz (27.7%), respectively.The measured dimensions for both the unperforated and perforated 3D printed antennas have been provided in The simulated and measured radiation pattern results for the 3D-printed SBF antenna are shown in Comparing the measured peak gain for the 3D-printed SBF antenna (15 dBi) to that of the aluminum prototype, which was measured to be 15.7 dB ,26,27. H−6 Ω·m) and superior EMI/RFI shielding, two perforated SBF antenna prototypes were 3D printed. The fabricated perforated 3D-printed SBF antennas are shown in For the fabrication of the perforated SBF antenna, a perforation radius of 4.5 mm was used. The PLA thermoplastic filament was the preferred material for the perforated antenna, as printing with ABS plastic resulted in many deformities and much stringing around the perforated areas. Stringing occurs when tiny plastic strings are left on a 3D-printed model. This happens mostly when the extruder temperature is too high, since it can cause the filament to overheat and ooze out of the nozzle. In the case of an ABS filament that has a higher melting point of 200 °C, this can be particularly problematic. To reduce the stringing problem, PLA with a lower melting point of 173 °C was used, and it produced a better finish. We also decided to compare the performance of two different metal paints: silver (Ag) and nickel (Ni) conductive paints. To compare the performance of the nickel conductive paint with the silver conductive paint with a lower resistivity (1.2 × 1011) results for both the nickel- and silver-coated protypes are given in 11 values for the nickel and silver prototypes were obtained using SR values of 1 Ω/sq and 0.1 Ω/sq, respectively. The measured S11 values of the nickel-coated and silver antennas were in very close agreement, as they had the same impedance bandwidth of 1.52 GHz (31.5%). This showed that using the silver paint instead of the nickel paint did not significantly affect the reflection coefficient of the perforated SBF antenna.The two antenna prototypes were connected to the waveguide feed and measured using the network analyzer. The measured and simulated  was fabricated. The simulated and measured results of the 3D-printed antenna were in close agreement with the results of the aluminum prototype for both S11 and radiation patterns. There was a 0.7 dB drop in the measured gain for the 3D-printed antenna compared to the aluminum prototype. This gain loss can be attributed to surface roughness and the lower conductivity of the metal coating. Employing the additive manufacturing technique in the design of the SBF antenna led to a significant reduction in the weight of approximately 70%.To further reduce the mass of the antenna, holes were introduced into the 3D-printed SBF antenna structure, resulting in a perforated SBF antenna design. From the simulation studies, we observed that the performance of the antenna did not change significantly with perforations that were much smaller than a wavelength. Two prototypes of this design were fabricated and coated using silver and nickel metal paints to compare their performances. The antennas were tested, and all performance metrics were achieved at 5.5 GHz. The silver paint was found to increase the gain of the perforated SBF antenna from 14.1 dBi (gain obtained for the nickel coating) to 15.2 dBi (1.1 dB increase in gain). While the silver paint increased the gain of the antenna, the financial cost was about 10 times the cost of the nickel paint for the same quantity. Therefore, the nickel paint was more cost effective than the silver paint. According to the results obtained, using both the perforation and 3D-printing techniques reduced the mass of the antenna by 80% while still meeting the design requirements, with a minimal gain loss of 0.5 dB .In future work, we aim to address the surface roughness issues by using LiDAR to measure and evaluate the precise surface roughness values of the 3D-printed surface and investigate the use of chemicals such as acetone for polishing the antenna. We also plan to conduct more research on advanced metallization techniques such as electroplating and jet metal processing. By reducing the surface roughness introduced by the 3D-printing process and obtaining uniform and precise metal deposition, we hope to further improve the gain of the 3D-printed antenna and get the performance as close as possible to that of the aluminum prototype. We also plan to carry out more optimization studies on the perforated antenna design, such as optimally packing the circular holes in a lattice-type structure and investigating combinations of multiple perforation sizes.


This paper presents a multiple-input-multiple-output (MIMO) antenna array with low-profile and flexible characteristics. Multiple microstrip patches are arranged in the E-plane configuration and decoupled by shorted quarter-wavelength stubs. The antenna has a small element spacing of 0.032 λ, where λ is a free-space wavelength at the center frequency. To demonstrate the feasibility of the proposed concept, a 1 × 4 MIMO array prototype is fabricated. The measured results on the fabricated prototype demonstrate that the MIMO antenna has good operation features at 4.8 GHz with a reflection coefficient of less than –10 dB and an isolation of better than 20 dB. Besides, good radiation patterns and broadside gain of around 4.5 dBi are also attained. The antenna also works in the bending mode and has the capability of extending to large-scale MIMO arrays. Such attractive features prove the utility of the proposed antenna in various modern electronic devices. Multiple-input-multiple-output (MIMO) antenna has achieved extensive research as an effective solution for 5G wireless communication services since it can increase the channel capacity without additional spectrum and alleviate the multipath fading problem , 2. AmonThere are various decoupling structures reported in the literature for mutual coupling reduction. They can be roughly divided into four categories. The first type is the counteraction scheme, which creates additional coupling paths between the MIMO elements to counteract the original coupling paths. This can be achieved with the aid of neutralization lines , 6, paraIt is worth noting that although high isolation can be achieved, most of the above-mentioned MIMO antennas are not suitable for flexible wireless systems, which are rapidly increasing. Nowadays, flexible antennas are in great demand for on-body and off-body communication, wearable sensing systems, radio frequency identification (RFID), and wireless body area networks (WBAN) , 22. TheIn this paper, a low-profile and flexible MIMO antenna is presented for operation at the N79-5G frequency band. The MIMO array consists of multiple microstrip patch elements arranged in the E-plane configuration. For mutual coupling reduction, the shorted quarter-wavelength stubs are inserted between the radiating patches. This contributes to producing a perpendicular mode on the non-exited element, rather than a similar operating mode as the excited element. The antenna is first designed and investigated with a two-element array. Then, further increment in the number of MIMO elements is considered. Finally, a four-element MIMO antenna array is fabricated and tested to validate the design concept. The measurements demonstrate the capability of working in both normal and bending modes of the proposed antenna. In comparison with the reported decoupling networks for microstrip patch antenna, the proposed decoupling structure has several advantages including high isolation improvement with small element spacing, applicable for multi-element MIMO array as well as its flexible capability.Ω SMA connectors are connected to the patches. To suppress the mutual coupling, a quarter-wavelength shorted stub is located between the MIMO elements. The antenna is designed on the ROGER-5880 substrate with a dielectric constant of 2.2. The optimized dimensions are as follows: Ls = 80, Ws = 60, H = 0.5, Lp = 20.3, Wp = 20.6, lf = 3.6, d = 2.0, rv = 0.2, l = 12.0, w = 0.8 (unit: mm).Assume that there are two antennas working in transmitting and receiving modes . If the TM01 is the dominant mode on the non-excited patch. Accordingly, the MIMO array with the shorted stub will perform better isolation than the other. A further demonstration can be observed in According to this theory, the mutual coupling in the MIMO system can be suppressed if the MIMO elements work in orthogonal modes. The optimization process of the proposed 2-element MIMO antenna can be divided into three main steps: operating frequency, impedance matching, and isolation. In general, these performance characteristics can be controlled independently.First of all, the resonance frequency of the proposed MIMO antenna can be controlled by tuning the length of the microstrip patch, Lp. Next, the matching performance is considered. The simulated S-parameter results for different values of lf are illustrated in l and rv. It can be seen that the effect of these parameters on the matching performance is minor. The |S11| resonance is stable around 4.8 GHz. In contrast, the isolation value is considerably affected. In the |S21| profile, the lowest |S21| peak shifts downwards with the increase of l and the decrease of rv. With proper values of l and rv, the isolation at 4.8 GHz will achieve the best performance.Finally, the isolation optimization is investigated. Figs Various decoupling structures have been reported for patch antenna arrays. However, these proposals are merely capable of alleviating mutual coupling between two adjacent elements. They might not work with multiple-port MIMO systems. In view of this point, a four-element patch array is further designed to demonstrate the decoupling capacity of the proposed decoupling structure in the MIMO patch antenna.According to the investigation on 2- and 4-element MIMO antenna, the design procedure for an arbitrary N-element MIMO array can be summarized as follows:Step 0: Choose the proper substrate materials that are thin and flexible.Two microstrip patches are arranged in the E-plane coupled configuration;Lp) is chosen about half-effective wavelength at the desired frequency band;The length of the patch (lf) is within the patch (lf < Lp/2);The feeding position  as the decoupling network is positioned between the MIMO elements;The ground stub . For the far-field test, when one port is excited, the other is terminated with a 50-Ω load. As observed, the simulated and measured gain values are almost similar. The antenna has a maximum gain of 4.7 dBi within the operating band from 4.75 to 4.83 GHz. The radiation patterns at 4.8 GHz of the proposed antenna are plotted in Due to the symmetrical antenna geometry, the far-field performances are only characterized by Port-1 excitation. i, j are port numbers, and R, T are the reflection and transmission coefficients. Ω is the solid angle of the far-field radiation patterns The MIMO performances of the proposed MIMO array are evaluated through several important parameters including the envelope correlation coefficient (ECC) and the diversity gain (DG). The ECC and DG are calculated based on the S-parameter Eqs The comprehensive comparison among microstrip patch MIMO antenna arrays is summarized and given in The proposed MIMO antenna is modeled on the thin substrate, the proposed antennas are expected to work effectively when mounting on curved surfaces. For measurement shown in The measured S-parameter results of the fabricated MIMO array in bending conditions are presented in This paper presents a flexible MIMO antenna array with multiple ports. An array of microstrip patch antennas are arranged in the E-plane configuration with edge-to-edge element spacing of 0.032λ. In order to suppress the mutual coupling, shorted quarter-wavelength stubs are positioned between the MIMO elements. A 1 × 4 MIMO array prototype is fabricated and tested to validate the feasibility of the proposed concept. The measured operating BW of about 1.6% with isolation of greater than 15 dB. At the center frequency of 4.8 GHz, the isolations among MIMO elements are always better than 20 dB. The MIMO array has also been demonstrated to work effectively in the bending mode. The proposed antenna can be a potential candidate in various modern electronic devices working in the 4.8 GHz N79-5G frequency band.


Hence, in total, the design incorporates four pairs of horizontally and vertically polarized resonators. The elements have compact profiles and resonate at 3.6 GHz, the main candidate bands of the sub-6 GHz 5G spectrum. In addition, despite the absence of decoupling structures, adjacent elements demonstrate high isolation. To the best of the authors’ knowledge, it is the first type of smartphone antenna design using dual-polarized self-complementary antennas that could possess anti-interference and diversity properties. In addition to exhibiting desirable radiation coverage, the presented smartphone antenna also supports dual polarizations on different sides of the printed circuit board (PCB). It also exhibits good isolation, high-gain patterns, improved radiation coverage, low ECC/TARC, and sufficient channel capacity. The introduced antenna design was manufactured on a standard smartphone board and its main characteristics were experimentally measured. Simulations and measurement results are generally in good agreement with each other. Moreover, the presented antenna system delivers low SAR with adequate efficiency when it comes to the appearance of the user. Hence, the design could be adapted to 5G hand-portable devices. As an additional feature, a new ultra-compact phased array millimeter-wave antenna with super-wide bandwidth and end-fire radiation is being introduced for integration into the MIMO antenna systems. As a result, the proposed antenna system design with improved radiation and multi-standard operation is a good candidate for future multi-mode 5G cellular applications.This paper introduces a multi-input multiple-output (MIMO) antenna array system that provides improved radiation diversity for multi-standard/multi-mode 5G communications. The introduced MIMO design contains four pairs of miniaturized self-complementary antennas (SCAs) fed by pairs of independently coupled structures which are symmetrically located at the edge corners of the smartphone mainboard with an overall size of 75 × 150 (mm In order to acquire the main themes of 5G networks, MIMO systems with an increased number of radiation elements must be considered for future wireless networks3. MIMO technology with multiple antennas can significantly amend the reliability function5. It has been extensively used in 4G LTE and is expected to be widely used in 5G. MIMO technology not only can significantly improve the system reliability but also increase channel capacity without requiring extra power at both the transmitter and receiver ends7. The use of diversity schemes in MIMO antenna configuration is also considered to be a crucial component of combating fading and enhancing the reliability of wireless links by sending the same signals with uncorrelated antennas8.The current generation (4G) wireless cellular systems are unable to meet future wireless communications requirements for high data rates. For these reasons, the 5th generation (5G) of wireless communications or mobile networks has been developed to address these challenges. It offers a variety of enhanced services for the internet of things (IoT), machine-to-machine (M2M), mobile broadband, massive MIMO, and ultra-reliable communications10. In addition, due to the limited available space on these devices such as smartphone boards, low-cost and compact planar microstrip antennas are an appropriate choice for cellular applications11. For sub 6 GHz 5G applications There have been many developments in smartphone antennas in recent years21. However, either these antenna arrays use resonators with single polarization, or they take up significant space on the mainboard. As smartphone PCBs are restricted in terms of antenna size, self-complementary antennas (SCAs) could be suitable for use in antennas, due to their compact size, simple structure, and ease of integration23. In addition, the SCA results in the geometry remaining unchanged when the metal and slot spaces are switched in a planar antenna24. We present here a new eight-port/four-antenna antenna array with miniaturized SCA radiators. In the single-element, the dual-polarized radiator has a self-complimentary structure and has been fed by independent coupled coaxial probes. The elements are highly miniaturized and resonate at 3.6 GHz. CST software package is used to design the antenna system25. Several unique characteristics distinguish this antenna from those reported in the literature, such as good isolation, high-gain patterns, excellent radiation coverage, low ECC/TARC, and sufficient channel capacity. Additionally, the proposed MIMO design is implemented, and its characteristics are analyzed. In order to verify the accuracy of the designed antenna performance, measurement results were carried out and the results were compared to electromagnetic simulations. Detailed descriptions of the double-fed SC resonator design and its array design are given.In hand-portable smart devices, high-efficiency and low-profile antennas that offer sufficient bandwidth and mutual coupling characteristics are very suitable26. In MM-Wave communications, phased array antennas with beam-steerable radiations are highly desirable since they enhance the radiation and connectivity of the systems. For smartphones, compact antennas can be used to form a linear phased array with high gain and directional radiation beams on the edges of the PCBs28. Moreover, end-fire antennas are more suitable for achieving the required full radiation coverage than conventional antennas, such as patch, slot, or monopole antennas29. Therefore, in addition to the proposed 3.6 GHz MIMO antenna, we have proposed a new mm-wave antenna package to operate at MM-Wave frequencies. The array consists of eight loop resonators arranged in a linear pattern, which can easily be integrated into smartphone antennas. The following sections present the design details, single-element performance, fundamental characteristics of the MIMO antenna system, and the suggested MM-Wave phased array, respectively.Apart from sub 6 GHz frequencies, the MM-Wave spectrum is also expected to be supported by 5G smartphones2. Figure 2: its parameters (in mm) are listed in Table Figure The double-fed antenna element is depicted in Fig. 30. The antenna configuration is flexible and different feeding techniques such as aperture coupling, microstrip line, and coaxial feedings can be applied. However, based on simulations, the employed feeding technique, which combines the coaxial and coupling feeds, offers a better performance in terms of impedance matching, bandwidth, and isolation. Moreover, since the elements are placed at the corners, the employed feeding technique is more appropriate to reduce the occupied space in the mainboard31.Figure L2). The S11 results of the dual-polarized SCA with various sizes of L2 are shown in Fig. 11 results of the SCA with various values of Lf are represented in Fig. 11 results of changing substrate thickness (h): increasing the substrate thickness allows good impedance matching with a wider bandwidth. The proposed design has also the potential to be used for 4G applications. The S-parameters for different types of substrates are shown in Fig. 32. A prototype antenna has been fabricated and tested. The fabricated sample and its measured S parameters are shown in Fig. The frequency response of the SCA can be easily changed and tuned to the desired band, by changing the design parameters. An important parameter to consider relates to the arm of the petal-ring structure  at 3.6 GHz. In addition, as shown in Fig. nm). Based on the simulation, sufficient efficiency properties were obtained, as shown in Fig. The S-parameters for the antenna array shown in Fig. 34. To have a better view, an illustration of the radiation patterns and gain values of the elements are given in Fig. 35. Therefore, the MIMO array design could be robust to the various holding positions of 5G smartphones.The broadside radiations of a single-element radiator with different polarizations are plotted in Fig. nn/Smn): well-defined results are obtained for SCA resonators. Additionally, the measurements are in good agreement with the simulations, indicating sufficient − 10 dB bandwidth and low couplings. A very slight variation was observed, possibly due to the errors in prototyping, feeding of antennas, and experimental setup. In addition, the S21 characteristics differ slightly from single antennas because of the large ground plane of the main design.A prototype of the proposed MIMO design has been developed and tested. The photos of the prototype sample (front/back layers with connectors) are shown in Fig. As the antenna pair’s performances were identical, radiations from the adjacent SCA radiators (Antennas 1 and 2) at 3.6 GHz were measured and plotted in Fig. 36. It is worth mentioning that these parameters have been computed using the below formula:ECC (envelope correlation coefficient) and TARC  properties of the presented array are examined in order to verify its capability in MIMO operation and considered in the following37. In addition, for further investigation of the MIMO performance across the band of interest, the computed channel capacity (CC) is also studied38. The CCL and CC are defined as follows:scale is the channel matrix.Figures Figures 39. Based on the results shown, the proposed antenna design exhibits adequate performance. A careful investigation reveals that the resonators partially surrounded by the hand phantom suffer the greatest radiation losses. In most cases, this is due to the nature of the tissue, since it usually absorbs radiation power41. The specific absorption rate (SAR) distribution for the elements (including Antennas 3 and 7) is investigated and shown in Fig. This Part of this study discusses the user effects as it pertains to antenna performance and SAR levels in appearances of hands and head phantoms. Figure nn, reflection coefficient (S11 to S88) results of the MIMO smartphone array structure in the presence of various integrated components  are represented in Fig. The SThe performance comparison between the introduced MIMO antenna and the reported design in the literature is represented in Table f = 8.7, Lf = 0.8, W1 = 1, L1 = 0.85, W2 = 3, L2 = 8, W3 = 0.85, d = 0.6. The schematic for the modified design is shown in Fig. nn and better than − 10 dB Smn at 3.6 GHz, which is the 5G band. However, as seen in Fig. 42.This section examines the performance of dual-polarized MIMO antennas with a full ground (GND). Modifying the configurations and increasing the sizes of dual-polarized patch radiators without complementary slots in the ground plane enables the proposed smartphone antenna to operate at 3.6 GHz. In this case, it is necessary to modify the parameter values of a single-element patch-ring resonator as follows (in mm): W = 14.5, L = 14.5, Wa × La which can be implemented in the same Rogers substrate material. The configuration of the single-element and its frequency bandwidth are shown in Fig. nn (S11 to S88) and Smn (S21 to S81) have been provided in Fig. In this section, a low-profile MM-Wave phased array with super wideband function is suggested to be incorporated in a shared board of the discussed MIMO smartphone antenna system. It contains eight end-fire loop resonators with a very compact size of WThe simulated efficiency results of the single element and the phased array are represented in Fig. Figure 43. Therefore, the suggested phased array can be used in various portable devices because of its numerous promising features, including highly miniaturized profile, super-wide bandwidth, well-defined end-fire radiation, wide beam steering capability, as well as sufficient efficiency/gain levels.The 3D radiation behaviour of the suggested phased array placed at the top and side edges of the smartphone mainboard with at 30 GHz are represented in Figs. An eight-resonator MIMO array design formed by employing dual-polarized/self-complementary structures is reported, at the 3.6 GHz 5G band. It is simply constructed on a smartphone board but meanwhile realizes satisfactory properties. Suitable input-impedance characteristics, mutual coupling, and diversity radiations are achieved. As compared with recently reported designs, the proposed smartphone antenna provides improved radiation coverage, less ECC/TARC results, and higher gain/efficiency characteristics. Additionally, it has a planar structure without any ohmic losses, making it an ideal candidate for 5G handheld devices. Meanwhile, the calculated channel capacity and its loss is about 40 and 0.5 bps/Hz for the desired frequency. Simulation and experimental results showed quite good agreement. By appropriately placing the proposed miniaturized resonators, high gain levels and pattern diversity can be achieved. Moreover, a new super-wideband/low-profile MM-wave phased array is suggested to be incorporated in a shared board of the smartphone antenna system. Its critical parameters  have been examined and sufficient results have been obtained. The suggested MIMO antenna systems can be used in multi-mode/multi-standard 5G cellular communications.


In this research, a multiband patch antenna design based on the superposition of multiple modes at the same resonant frequency is presented. The concept of the contribution of lower order modes (LOMs) with the higher order dominant modes (HODMs) is investigated using characteristic mode analysis (CMA). In order to provide similar broadside radiations in three bands, the radiating capability of the LOMs is enhanced in the resonant frequencies of HODMs. These HODMs when excited alone provide null in the broadside radiations of the antenna. Using a single feed, enhancement in the broadside radiations of the antenna is achieved with the superposition of multiple modes at the same resonant frequency. Based on the proposed concept two antennas have been designed and fabricated. The antenna provides stable and enhanced radiations. simulated and measured results are in good agreement. These antennas find application in many applications including communication systems, base station antennas, 5G communications, satellite communication etc. Modern communication devices need an antenna to be low-cost, multiband, and smaller in size. Various techniques to achieve multiband antenna operation have been reported in the literature, where additional modes within a band of interest are introduced by using: fractals10, split ring resonators5, U-slots, vias and varactor diodes14, use of dual-feed and more than one radiating element14, slot meandered patch15 and V-shaped antenna configuration16, mobile handset antennas18, metamaterial loaded antenna19, and slotted ground20. Patch antenna provides attributes of low cost, compactness, and conformability, making it a suitable candidate for modern antenna systems. However, the patch antenna suffers from the nulls in the radiations and grating lobes due to the undesired radiations offered by the higher order modes24, consequently making it very difficult to achieve desirable radiations in multiple bands of an antenna. Multiband and dual-band antennas with nulls in the radiations cannot be used in applications where broadside radiations are required in all bands26. To achieve broadside radiations in the second band, slots are incorporated along the radiating edges of the rectangular patch to perturb 21. A slight dip in the radiations of 24, is removed by introducing slots in22 which provided current perturbations, thereby making perturbed 27. Higher order 28. Patch and monopole modes in a shared aperture are also used to excite two modes for dual-band operation29.With the advent of modern wireless communication32. This attribute makes it an excellent antenna analysis tool to identify radiating modes inherent to the structure. Various efforts have been made by the researchers to design dual-band and triple-band antennas using characteristic mode analysis (CMA), including multiband MIMO antennas36, fractal shaped dual-band antenna37, U-slot loaded triple-band antenna13, loading triangular slot38, vias39 and metasurfaces39 to introduce additional modes within a band of interest.Lately, the theory of characteristic modes (TCM) gained popularity in solving antenna design problems. TCM provides complete information about radiating modes of an arbitrarily shaped PEC structure in the range of frequencies under consideration41 is based upon the removal of higher order modes (HOMs) using CMA to enhance antenna radiations. In40 an unwanted HOM is suppressed to enhance the realized gain of the antenna by using a metasurface in front of the antenna array, in41 suppression of HOMs is carried out by inserting vias and slots in the unit cells of a metasurface structure to improve performance of the antenna in terms of radiations. Nevertheless, very few publications are related to the pattern enhancement by manipulation of HOMs in CMA. To the author’s knowledge, none of the published work is related to the suppression or contribution of LOMs along with DMs to enhance antenna radiations. This work presents the intentional excitation and superposition of multiple LOMs with the HODMs to remove radiation nulls in the higher bands. The significance of the LOMs is enhanced beyond their resonant frequencies, and the superposition of non-significant LOMs with the HODMs at the same frequency is carried out to enhance antenna radiations. It is found that the excitation of multiple modes at the same resonance provides more control over shaping the antenna’s radiation characteristics. The proposed technique is applied to the Partial Koch fractal structure reported in42, which suffers from the null in the radiations in the second band. Current authors introduced pi-shaped slots on the surface of the structure as reported in43, however, perturbations produced by the slots only removed radiations nulls in the H-plane of the second band. To show the problem of radiation nulls, and provide the superiority of the proposed technique, a multiband antenna using the excitation of DMs in their respective bands is also presented.For the multiband antenna operation using CMA, multiple radiating characteristic modes (CMs) are required to be excited in the band of interest. The number of CMs is directly proportional to the size of a radiator; hence density of modes increases with the increase in frequency. The radiation patterns of an antenna are shaped by the radiations of DMs, consequently, for a multiband antenna with usable broadside radiations, dominant modes (DMs) within the band should provide usable radiations. Furthermore, coupling of undesired modes with the source also deteriorates the radiation performance of the antenna in higher bands. Some of the latest reported research work by Chen et al.32. Eigenvalue is given by:30, as given byTCM provides complete information about the resonant behavior and radiating mechanism of an arbitrary shaped conducting structure. Deep physical insight into the radiating mechanism is provided in terms of modal currents and parameters such as eigenvalue 31, represents the contribution of a mode to the total antenna radiations when an external source is applied, given by:13. Each mode at its resonant frequency contributes maximum towards the total current and radiations of the antenna and is called DM. The Resonant frequency of the mode is determined from the MS graph, a mode is resonant and called a DM at the MS peak. 31.29 are given by:Closeness of the modes to the resonant frequencies in the limit defined by CMA values Amount of the coupling of the mode to the source.MSentclass1pt{minimaExcitation of only DMs for multiband antenna operation. In this approach, only one mode is excited in each band, so the currents and radiations of the antenna are shaped only by DM in each respective band.Excitation and superposition of multiple modes at the same resonant frequency. In this approach, multiple modes are excited, therefore resultant antenna currents and radiations are the superposition of LOMs and DMs that are excited simultaneously at the same resonant frequency (superposition-based approach).A flowchart explaining both methodologies is also provided.To highlight the performance benefit of the proposed technique, we have presented the antenna design using the two techniques. The following two approaches are outlined in this section A flowchart for the procedure to design a multiband antenna with the excitation of DMs is provided in Fig. A flowchart for the proposed technique which uses the superposition of DMs and LOMs is provided in Fig. 42 provided in Fig. 43, moreover it provides a large number of modes within the band. Simulations are performed with patch, lossless substrate  are shown in Fig. In this section, a design procedure based on the superposition of multiple modes at the same resonant frequency is provided, and as design example two antennas are presented. Currents of LOMs are combined with the HODMs to remove the current nulls, and superposition of subsequent radiations to provide enhancement in broadside radiations of the antenna in the second and third bands is achieved. Feed point location is identified using currents of both LOMs and HODMs. Antennas are designed by using the procedure outlined in Fig. Table To improve antenna performance in the second band, the superposition of LOMs and DMs as provided in Table Similarly, to improve antenna performance in the third band, superposition of mode 1, 3 & 5 of Antenna 1(b) and mode 2, 4 & 5 of Antenna 2(b) is carried out at 5.10 GHz as shown in Fig. To reveal the radiating mechanism of the antenna with excitation of multiple modes at the same resonant frequency, parametric analysis of the Partial Koch antenna at different air gap heights Progression of the radiating behavior of LOMs with the increase in air gap heights 2D rectangular modal directivity comparison graph of Antenna 1(b) & Antenna 2(b) is provided in Fig. To excite the required modes simultaneously at the same resonant frequency, a high current density region that is common among all DMs and LOMs is chosen as the feed point, and CMA of the antenna with the source is carried out in the frequency range of 2–7 GHz. Simulations are performed with infinite ground plane conditions, the substrate used is lossless Rogers Duroid 5870 with permittivity of 2.33, the thickness of the substrate used is 1.5 mm, and an air gap of 4.5 mm is used between the substrate and ground plane, and coaxial probe is used. From the MWCs in Fig. Parametric analysis of Antenna 1(b) and Antenna 2(b) is performed with the same conditions as provided before. The analysis includes optimization of reflection coefficients of both antennas in three bands. Simulated reflection coefficients of Antenna 1(b) and Antenna 2(b) at different air gap heights Fabricated prototypes of Antenna 1(b) and Antenna 2(b), and measured reflection coefficients are provided in Fig. A comparison of simulated 2D normalized gain patterns of Antenna 1(a) Fig.  and AnteTo compare the proposed technique with the other methods, a performance comparison is provided in Table A procedure is proposed to design antenna using MC of LOMs at the resonant frequencies of HODMs for multiband antenna operation with improved radiation performance and control. The radiating capability of modes having poor radiations is enhanced beyond their resonance to shape the broadside radiations of antenna in multiple bands. The carefully chosen feed point is identified to excite desired modes with the required contributions. As a proof of concept, this technique is utilized to design a multiband patch antenna with stable and enhanced radiations. The concept is verified by achieving stable and enhanced radiations in all bands, also null in the radiations of the second and third bands is successfully removed. Close agreement is observed between simulated and measured results. The proposed procedure proves that multiband behavior with improved radiations can be achieved by properly using the MC of LOMs in the resonant frequencies of HODMs. Required results are achieved without any modifications in the structure of the patch or designing complex feed structure.


This article presents a quad-element MIMO antenna designed for multiband operation. The prototype of the design is fabricated and utilizes a vector network analyzer (VNA-AV3672D) to measure the S-parameters. The proposed antenna is capable of operating across three broad frequency bands: 3–15.5 GHz, encompassing the C band (4–8 GHz), X band (8–12.4 GHz), and a significant portion of the Ku band (12.4–15.5 GHz). Additionally, it covers two mm-wave bands, specifically 26.4–34.3 GHz and 36.1–48.9 GHz, which corresponds to 86% of the Ka-band (27–40 GHz). To enhance its performance, the design incorporates a partial ground plane and a top patch featuring a dual-sided reverse 3-stage stair and a straight stick symmetrically placed at the bottom. The introduction of a defected ground structure (DGS) on the ground plane serves to provide a wideband response. The DGS on the ground plane plays a crucial role in improving the electromagnetic interaction between the grounding surface and the top patch, contributing to the wideband characteristics of the antenna. The dimensions of the proposed MIMO antenna are 31.7 mm × 31.7 mm × 1.6 mm. Furthermore, the article delves into the assessment of various performance metrics related to antenna diversity, such as ECC, DG, TARC, MEG, CCL, and channel capacity, with corresponding values of 0.11, 8.87 dB, −6.6 dB, ±3 dB, 0.32 bits/sec/Hz, and 18.44 bits/sec/Hz, respectively. Additionally, the equivalent circuit analysis of the MIMO system is explored in the article. It’s worth noting that the measured results exhibit a strong level of agreement with the simulated results, indicating the reliability of the proposed design. The MIMO antenna’s ability to exhibit multiband response, good diversity performance, and consistent channel capacity across various frequency bands renders it highly suitable for integration into multi-band wireless devices. The developed MIMO system should be applicable on n77/n78/n79 5G NR (3.3–5 GHz); WLAN (4.9–5.725 GHz); Wi-Fi (5.15–5.85 GHz); LTE5537.5 (5.15–5.925 GHz); WiMAX (5.25–5.85 GHz); WLAN (5.725–5.875 GHz); long-distance radio telecommunication ; satellite, radar, space communications and terrestrial broadband ; and various satellite communications . Multiple input multiple output (MIMO) antenna technology is an indispensable option for wireless transmissions to meet more data needs and increase spectral efficiency. MIMO antenna configurations for 4G will also be an important system for enabling 5G . This 5 This article consists of three sections. The first section covers the details of antenna geometry. The second section covers the detailed analysis of different diversity characteristics parameters like ECC, DG, The proposed antenna element has a dual-side three-stage stair and a straight stick on the top of the stair. The partial ground plane has a small rectangle microstrip line just beneath the feed line. The introduction of a DGS on the ground plane serves to provide a wideband response. The DGS on the ground plane plays a crucial role in improving the electromagnetic interaction between the grounding surface and the top patch, contributing to the wideband characteristics of the antenna . This DGThe size of the single-element antenna is 14.7 mm × 12 mm × 1.6 mm. In this design, the following design variables are considered, which are “a”, “b”, “c” and “d”. For optimized performance of the antenna, the design variables a, b, c, and d are considered as 1.5 mm, 2.1 mm, 3 mm, and 2.3 mm, respectively. The top and bottom views of the proposed antenna element are shown in Furthermore, the conceptual equivalent circuit model for the proposed antenna was developed a. The inThe real (resistive part) and imaginary (reactance part) parts of the input impedance of the proposed MIMO antenna system are depicted in The optimized values of lumped components are shown in The developed MIMO system should be applicable on n77/n78/n79 5G NR (3.3–5 GHz); WLAN (4.9–5.725 GHz); Wi-Fi (5.15–5.85 GHz); LTE5537.5 (5.15–5.925 GHz); WiMAX (5.25–5.85 GHz); WLAN (5.725–5.875 GHz); long-distance radio telecommunication ; satellite, radar, space communications and terrestrial broadband ; and various satellite communications . As mm-wave has a high carrier frequency, the signal baud rate could carry more data, resulting in a high data rate (bits/s/Hz) and a reduction in the buffering time in delivering high-quality videos . The mm-The developed MIMO antenna covers a 37–40 GHz frequency band, which has been reported in ,35,36. TThe interfacial current distributions at a frequency of 11.1 GHz, 31.6 GHz, 37.8 GHz, and 46.7 GHz are illustrated in L is the perimeter of the patch. In this article, Considering the ‘m’ number of null points that exist over the perimeter of the patch, then the guided wavelength , 26.4–34.3 GHz (% bandwidth of 32.45), and 36.1–48.9 GHz (% bandwidth of 32.74). Over the entire frequency bands in OFB1, OFB2, and OFB3 the simulated minimum values of port isolation are 15.7 dB, 23.2 dB, and 23.2 dB, respectively; however, the measured minimum values of port isolation are 15 dB, 22 dB, and 22.3 dB, respectively. The simulated and measured S-parameter is depicted in i’ is excited, and j’ is excited. The solid angle is denoted by the symbol Ω. The value of the ECC that is closest to zero implies that the MIMO antenna system has strong isolation and excellent diversity gain. Equation (5)  is used The degree of the improvement of the MIMO system is determined by diversity gain (DG). ECC can acquire the DG  calculatThe simulated and measured ECC and DG over the frequency bands OFB1, OFB2, and OFB3 are depicted in The simulated ECC and DG over the frequency bands OFB1, OFB2, and OFB3 are better than , , and , respectively, whereas the measured ECC and DG over the frequency bands OFB1, OFB2, and OFB3 are better than , , , respectively.XZ plane. At the frequency 11.1 GHz, 31.6 GHz, 37.8 GHz, and 46.7 GHz, the co-polar major lobe direction of Ant1-Ant4 are XZ plane. Such complementary radiation pattern qualities enable a low value of ECC between antennas and are also useful in producing self-isolated MIMO antenna systems. In Based on the observations, it is discovered that the radiation patterns of  and  are complementary to each other in the The mean effective gain (MEG) is a key measure for characterizing the performance of MIMO diversity. It compares the power received by the isotropic antenna to the power gained by the diversity antenna in a fade environment. The MEG of tion (7) .(7)MEGiThe typical value of TARC is characterized as the square root of the overall reflected power divided by overall incident power. TARC provides a valid estimation of MIMO antenna efficiency since it contains information about the mutual coupling effect. The TARC for the N unit MIMO antenna setup can be computed using Equation (8) .(8)TARCThe value of TARC is calculated with the assumption of An additional fundamental diversity attribute measure that may be used to assess MIMO performance is CCL, which shows the greatest rate at which data can be delivered without experiencing significant losses. Equation (9) ,50 can bErgodic channel capacity (CC), which is determined under the presumption of zero ECC among the transmitting antennas, is the most efficient diversity characteristics metric for a MIMO system. Equation (10) is used to evaluate the channel matrix “H”.ained in .Further the The variables used in Equation (11) are ion (12) ,53 is usHere, the variables in Equation (12), The ideal CC for 4 × 4 MIMO and 2 × 2 MIMO are 22.2 bps/Hz and 11.3 bps/Hz at SNR = 20 dB, respectively; whereas, for 4 × 4 MIMO and 2 × 2 MIMO, the ideal CC is 10.94 bps/Hz and 5.53 bps/Hz at SNR = 10 dB, respectively. The CC should be at least 60% of the ideal value for a MIMO antenna to work well. The simulated CC over the operating frequency bands OFB1, OFB2, and OFB3 is 4.53–8.95 bps/Hz, 7.46–8.70 bps/Hz, and 7.20–8.18 bps/Hz, respectively, at SNR = 10 dB, whereas the CC at SNR = 20 dB is 12.60–19.57 bps/Hz, 17.46–19.21 bps/Hz, and 17.20–18.56 bps/Hz, respectively. The measured CC over the operating frequency bands OFB1, OFB2, and OFB3 is 4.35–8.62 bps/Hz, 6.89–8.68 bps/Hz, 7.02–8.01 bps/Hz, respectively, at SNR = 10 dB; whereas, the CC at SNR = 20 dB is 12.13–19.13 bps/Hz, 16.55–19.18 bps/Hz, and 16.92–18.44 bps/Hz, respectively. The channel capacity versus frequency plot is depicted in Since the proposed antenna is wideband in nature, it is important to check its time domain characterization. Important time domain characteristics like group delay, phase response, and isolation analysis by keeping the antenna in face-to-face (FTF) and side-to-side (STS) orientation are studied. The two similar antennas are placed in two different configurations maintaining 300 mm, which are depicted in In order to discuss the time-domain analysis, the fidelity factor of antenna and system fidelity factor play a crucial role. The antenna fidelity factor (FF) is obtained by calculating the cross-correlation of the radiated E-field and the input signal; however, the system fidelity factor (SFF) is calculated by the cross-correlation between the transmitted pulse and the received pulse. The simulated value of FF and SFF are calculated in two different orientations (face to face and side by side), by keeping the proposed antenna 300 mm apart. Here the Gaussian sine pulse is given as input to the Tx port in order to find the FF and SFF. The obtained value of FF and SFF is presented in The antenna’s compact dimensions impact its transient behavior, and the level of pulse distortion is considered acceptable , whiThe proposed design operates on three wide bands that cover bands 3–15.5 GHz and two mm-wave bands . The minimum port isolation and ECC over the bands 3–15.5 GHz, 26.4–34.3 GHz, and 36.1–48.9 GHz are observed as ,  and , respectively. The diversity performance of the MIMO antenna is performed by analyzing diversity gain, average efficiency, total interference ratio, channel capacity loss, and ergodic channel capacity. The MEG and TARC in all operating bands  are less than 0.2 dB and −4.1 dB, respectively. The CCL in all operating bands is less than 0.3 bps/Hz and the range of CC in operating bands 3–15.5 GHz, 26.4–34.3 GHz, and 36.1–48.9 GHz are 12.13–19.13 bps/Hz, 16.55–19.18 bps/Hz, and 16.92–18.44 bps/Hz, respectively. The simulated time-domain analysis in two different orientations, which are face to face (FTF) and side to side (STS), are studied. In both orientations, the group delays (GD) lie between 1010 and 1085 ps, and the S21 magnitude is less than −75 dB in all operating bands. The S21 phase provides linear characterization in all operating bands. The simulated values of FF and SFF are  and  in FTF and STS orientations, respectively. In all operating bands, the measured values provide good agreement with respect to the simulation ones, which indicates a good candidate for operation in all investigated bands.


This article investigates the inverse design of a reconfigurable multi-band patch antenna based on graphene for terahertz applications to operate frequency range (2–5THz). In the first step, this article evaluates the dependence of the antenna radiation characteristics on its geometric parameters and the graphene properties. The simulation results show that it is possible to achieve up to 8.8 dB gain, 13 frequency bands, and 360 In wireless communication, the need for multi-band antennas has increased due to a reduction in the number of antennas, a reduction in the complexity and cost of the system, and providing the possibility of integration with other circuits of the structure5.Nowadays, the terahertz band is used in wireless telecommunications6, patch antennas9 ultra-broadband absorbers10, and photoconductive antennas11 has been reported. In12, a dual-band antenna with an average gain of 2.45 dB is designed by creating two circle strips on the graphene. In13, a three-band frequency reconfigurable antenna has been proposed for a slotted patch graphene antenna. In14, a three-band antenna is implemented with a series feed circle graphene patch with a gain close to 10 dB. In15, a four-band antenna has been reported for a four L-shaped stub graphene patch antenna. Generally, for graphene antennas, it is possible to change the number of operating frequency bands by the variation of graphene chemical potential, similar to17 in which a four-band and three-band graphene antenna has been designed with the gain of 2.58 dB and 9.51 dB, respectively.On the other hand, the use of graphene has been very impressive in recent years in the field of Nano-electronic and THz devices due to its high conductivity and the changeability of the conductivity by tuning the bias voltage. The use of graphene in THz imaging19, design of metasurfaces22 and metamaterials25, design of photonic structures27,beamforming29, design of antennas32.Making less computational time of resources with an acceptable result is of substantial importance in electromagnetic applications. In this regard, the machine learning approach has recently demonstrated outstanding performance compared to the computational and iterative methods in dealing with electromagnetic problems. Deep learning (DL) is a subset of machine learning (ML) with more robust computing capabilities, which is based on neural networks (NNs) and can learn the nexus between inputs and outputs. After learning, the designed model based on trained data can show a reasonable prediction as outputs for various given inputs in a fraction of a second. By taking advantage of this, DL was a suitable technique for inverse scattering problems33. Usually, these DNN architectures include multiple networks to handle a specific part of the problem and have high structural complexity. Some of these techniques are used to solve the one-to-many issue. The one-to-many issue is a challenging problem in machine learning, which refers to a situation in which a single input is associated with multiple outputs. A tandem architecture is presented in34 by bringing together forward modeling and inverse design. The approach demonstrated in this model overcomes the one-to-many issue better than the inverse design model for the designing of nanophotonic structures. Then to overcome the limited generalization ability of23, a probabilistic graphic model introduced as an all-inclusive explanation for metamaterial design.The deep neural network (DNN) architecture has been evolving over the years with new techniques and advancements. We can mention some recent techniques that applied in electromagnetic fields35. Second, by giving the antenna dimensions as input, the antenna radiation will be estimated as output in real-time so DL and ML can speed up the antenna simulation directly31. And third, the most widely used method is the inverse design of the antenna using DL. The required radiation pattern characteristics are provided as input, and the DNN’s output estimates the antenna parameters. In this case, depending on the circumstances, the antenna parameters may be fully adjustable or non-adjustable32. Although in29 a We have classified using the neural networks in antenna design into three approaches. First, NNs and ML enhance some radiation properties by optimizing the antenna parameters and can not control antenna radiation patterns in real-timeIn this article, an inverse design of reconfigurable graphene circular patch antenna at THz frequencies is proposed and surveyed to the realization of an intelligent antenna for 6G wireless communication. Also, for the first time we apply a chemical potential of graphene as a reconfigurable component in the output of DNN to control the radiation properties in real-time. The antenna parameters are divided into two groups, graphene and antenna parameters. After the analysis, we generated a data set and filtered it with two specific conditions. Then, a DNN model is presented, that can accurately predict the values of the graphene properties and antenna parameters for the desired S-parameters and radiation pattern of the antenna.For this, in Sect. “The flow chart of the activity steps is plotted in Fig. mentclass2pt{minimThe structure of the antenna along with its parameters is plotted in Fig. mentclass2pt{minimeV to 0.8 eV, the operation frequencies of the antenna have been shifted while the radiation pattern at the 2.34 THz frequency remains relatively constant. In this simulation, ps, eV to 1.3 eV, the return loss of the antenna remains relatively constant while the beam direction has been tilted about 30 degrees at the 3.21 THz frequency. In this simulation, ps, Due to the use of graphene in the patch and ground of the structure, a reconfigurable radiation pattern or a reconfigurable operating bandwidth is the inherent property of the structure. As shown in Fig. eV), relaxation time (ps), substrate thickness  has been used to calculate the difference between the real value and the value estimated by the model, as given in equation (The loss and accuracy diagrams of the proposed model have been sketched in Fig. For evaluation of the proposed model, we have provided two arbitrary samples whose specifications are given in Figure In this article, a planar graphene antenna was investigated and its radiation characteristics for different substrate thicknesses and graphene characteristics including chemical potential and relaxation time have been extracted by full-wave FEM simulations. Then, these parameters were examined as the design inputs and it was shown that in the proposed structure, by choosing the appropriate values for the input, the number of bands and their resonance frequencies, antenna gain and main lobe directions can be set. The simulations have been performed in the [2–5] THz frequency band and it has been shown that the maximum gain of 8.8dB and up to 13 frequency bands can be achieved. Due to the complexity of the design and in the following, a deep neural network has been used to provide a design solution. This network is trained based on the categorized simulation results and the network parameters are determined in such a way that the most accurate matching between the estimated and simulated parameters has been achieved for minimum input data. In the end, it has been shown that the optimal radiation parameters can be estimated with an error of less than 3%. This feature can be used to design antennas in various applications. In the simulations, we encountered one-to-many issues. In fact, the analysis of the simulation results shows that for the antenna parameters of [


The experimental investigations reveal that the reported design adequately retains its performance while operating in close proximity to human beings. The presented Specific Absorption Rate (SAR) analysis reveals 0.297 W/kg calculated at 0.5 W input power, which certifies that the proposed antenna is safe for use in wearable devices.A compact, conformal, all-textile wearable antenna is proposed in this paper for the 2.45 GHz ISM  band. The integrated design consists of a monopole radiator backed by a 2 × 1 Electromagnetic Band Gap (EBG) array, resulting in a small form factor suitable for wristband applications. An EBG unit cell is optimized to work in the desired operating band, the results of which are further explored to achieve bandwidth maximization via floating EBG ground. A monopole radiator is made to work in association with the EBG layer to produce the resonance in the ISM band with plausible radiation characteristics. The fabricated design is tested for free space performance analysis and subjected to human body loading. The proposed antenna design achieves bandwidth of 2.39 GHz to 2.54 GHz with a compact footprint of 35.4 × 82.4 mm The developments in body area network (BAN) devices have been fueled by the rise in the demand for IoT (Internet of Things) connected devices. Wearable devices have accelerated the digital transformation of several sectors, including health monitoring, tracking, rescue operations, and gaming ,2. WearaCurrent technological solutions are not entirely enough to fulfill the demands of pervasive health monitoring in coming era. There is a great demand to scale up the research towards reliable wireless connectivity, smart sensors, wearable devices, and wireless implants . With adWearable technology has the potential to revolutionize the lifestyle of the common man. Its tendency to simultaneously become faster and smaller holds out the prospect of a future orbiting around wearables. The trend of consumer preferences reveals an ever-increasing demand for wearable health monitoring devices. Users find self-tracking devices helpful in keeping them engaged with their health . HoweverDesigning high performance antennas is very simple, provided that ideal conditions are applied. A bulky antenna structure operating in an environment where no interferences exist is all that is needed. Efficient antenna design is obtained when the size is about half the wavelength. This means that an antenna of almost 8 cm is required for the 1900 MHz Global System for Mobile Communications (GSM) band. There should be no interfering electronics nearby when this antenna is operating. This is obviously not possible; hence, researchers continue working to optimize antennas in size and performance as per the required application.Wearable antennas have become more common in recent years due to their appealing qualities and potential to provide wireless environments and communication while being lightweight, tiny, affordable, and customized. Maintaining the uniformity of features in various environments is one of the main problems encountered in the wearable antenna development, covering aspects such as temperature, humidity, proximity to people and their clothing, wash cycles, etc. . BecauseSeveral articles have presented textile wearable antennas ,17,18 anDesigning antenna for on-body operation is challenging from various aspects. The antenna should be compact and conformal to offer a good user experience. Daily life activities should not be interrupted by wireless data transmission. Furthermore, human tissues have high permittivity, which allows for near-field coupling with the body . An anteA fabric antenna for wearable applications that is low-profile, small, conformal, and has HIS was presented in . BecauseThe development of the antenna design proposed in this work was carried out with the aim of obtaining a compact footprint and reduced geometrical complications to ensure good fabrication on textiles while ensuring that the design is able to offer maximum user comfort. These objectives were met by designing the integrated antenna on a denim substrate with a compact footprint. An EBG unit cell is set to operate in the ISM band, with a 2 × 1 EBG array is selected for the sake of a miniaturized configuration. When the number of unit cells is limited by size constraints, the usual approach of placing an EBG array behind an antenna operating at the same frequency ends with inferior performance in terms of bandwidth and radiation efficiency. For this reason, a novel approach utilizing a 2 × 1 metasurface with a floating ground plane is adopted. This approach allows for a more compact structure with enhanced performance. A similar kind of a wearable antenna was presented in  for the Because the ground plane of the EBG structure is displaced, the design methodology is quite different from the conventional approach , which bX1, Y2, and Y3. The resonance frequency is recorded in the zero-degree reflection phase, and is required to be set at 2.45 GHz. It is observed that by increasing or decreasing the values of the design variables the resonance frequency is moved to either a higher or lower spectrum. However, there is very little impact on the bandwidth. Therefore, the bandwidth offered by the unit cell is mainly controlled by the Uh parameter. The other design parameters control the shifting of the resonance frequency.With the objective of reducing the number of adjustable variables in the design, a simple slotted-square EBG unit cell was tuned to operate in the desired band using a denim substrate  and EBG ground plane displacement (d) is shown in d by 0.2 mm causes a frequency shift to a higher spectrum and reduces impedance matching. These values are optimized to achieve the desired resonance frequency with optimum band coverage.With the developed compact 2 × 1 EBG surface, a planar monopole antenna is placed above it at a close distance. The spacing allows for electromagnetic coupling; thus, the EBG surface plays a role in controlling the emitted radiations as well. The antenna dimensions are optimized to produce resonance at 2.45 GHz. The final integrated design with dimensions is presented in In this section, we present the simulation and experimental analysis of the proposed wearable antenna. The prototype development involves conductive fabric and readily available denim fabric, achieving an all-textile solution to wearable devices.The accuracy and manufacturing efficiency of low-cost wearable antenna designs are determined by fabrication procedures. Wet-etching, screen printing, ink-jet printing, and embroidery processes are the most widely used ways of wearable fabrication reported in the literature ,42,43. TThe fabricated antenna is shown in The antenna prototype was experimentally analyzed for the performance parameters of reflection coefficient shown in , gain s, and radFree space measurement results predict admirable performance of the proposed wearable antenna. However, it is important to analyze the performance variations in the wristband case. The fact that the human body consists of lossy tissues affects the antenna’s efficacy. The electrical properties of human organs are distinct, making them suitable EM (electromagnetic) energy absorbing materials. Thus, the radiated energy of the antenna is subject to being partially absorbed by the user’s body. The amount of this effect depends on several parameters, including the distance between the antenna and the user and the radiation characteristics of the antenna. The EM wave loses energy upon interaction with a dielectric medium. The human body is a non-homogenous dielectric medium that is characterized by the There are different ways to develop a human body model in a simulation tool to observe on-body performance. There are human anatomical models available in EM simulation tools, such as the voxel phantom model, which can accurately describe the human anatomy. These models are more rigorous and realistic than the multilayered phantom model; however, they require substantial computational resources. For analysis based on simulations, we constructed a multi-layered phantom model in the simulation tool. The dielectric constant, loss tangent, and mass density of each layer were defined as per the reported data from the literature. The model shown in The human body-loaded antenna performance parameters of the reflection coefficient (in decibel), gain, and radiation pattern are depicted in In addition to the above-mentioned analysis, the presented design operation was further investigated by mounting the antenna over the wrist of the body. The recorded results are presented in When the human body is exposed to EM radiation, one result is elevated body temperature . RegulatIn addition to the requirements of enhanced performance parameters of wearable antennas, arguably one of the most critical concerns for any device operating close to the human body is its impact on human health. This matter is assessed by SAR analysis, which indicates the amount of radiation absorbed by human tissues. The standard limitations are to be followed by all wearable devices. Thus, the SAR value should be less than 1.6 W/kg for 1 g of tissue, a limit set by the FCC. SAR dissemination is calculated for the diverse layers of human tissues. It is obvious from A comparison between the proposed wearable antenna and works published previously in the literature is provided in In this paper, we propose a novel flexible wearable antenna exploiting a floating-ground based EBG to enhance antenna performance. The achieved bandwidth is able to account for possible band shifting under bending or human body loading effects. The antenna was fabricated using Shieldex Nora Dell metallized fabric over a denim substrate layer. Cutting, pasting, and alignment of the design was carried out using A Silhouette Cameo machine. The computed overall far-field properties of the antenna indicate good performance with adequate gain and efficiency. An SAR analysis is presented, and the results validate the appropriateness of the suggested embedded design for wearable devices.