A miniaturized folded dipole patch antenna (FDPA) design for biomedical applications operating at sub 1 GHz (434 MHz) band is presented. Antenna is fabricated on FR-4 substrate material having dimensions of 16.40 mm  The application of biomedical antenna lies in early-stage cancer detection3, glucose monitoring4, Cardiac pacemaker5, endoscopy6 and cancer treatment through hyperthermia.Advanced development in antenna designing urges the need to integrate antenna with biomedical applications7. Hyperthermia is a non-invasive and effective cancer treatment technique followed by conventional Surgery, chemotherapy and radiotherapy, and it is considered an adjuvant technique for cancer treatment8. Hyperthermia has been used safely for treating previously treated tumors or regrowth of the tumor cells9 in which temperature is raised up to 45 10.Hyperthermia is an important application of bio-medical antennas where a certain temperature is applied for a limited duration to destroy cancer cells12, along with 902.8–928 MHz frequency band13. The Federal Communications Commission (FCC) recently revealed the wireless medical telemetry service (WMTS) band for frequencies between 608–614 MHz and 1.395–1.432 GHz14. Moreover, THz frequency band is also gaining popularity with antennas designed at 0.5 to 2 THz15, 1–4.43 THz16, 3.6–7.4 THz, 8.25–10.0 THz17, and 430 to 750 THz18.The frequency band used for the biomedical antenna are Medical implant communication system (MICS) at (401–406) MHz and Industrial, scientific and medical (ISM) bands at 434 MHz, 915 MHz and 2450 MHz19. So, 434 MHz is considered a suitable frequency band for hyperthermia11. However, the larger antenna size at such a lower frequency may limit its applications for developing compact, portable, or implantable devices, especially for endoscopy, where the antenna is implanted inside the capsule20. Therefore, the major challenge while designing the compact antenna at such a lower frequency is to miniaturize its size with suitable radiation performance22.Biomedical antennas operating in low-frequency bands are more suitable for cancer treatment through hyperthermia due to higher penetration depth of electromagnetic fields at low frequencies23. In addition, the performance of an antenna is also influenced by its proximity to the human body for hyperthermia applications24, hence, the appropriate design of miniaturized antenna with stable radiation performance near the human body is also a difficult task.Small antenna size also requires a complex matching network for impedance matching because of small input resistance and large reactance at the input. Therefore, the design of a simple impedance matching configuration for a miniaturized antenna is another challengeThere are three categories of clinical hyperthermia: local, regional and whole-body.In local hyperthermia, cancer cells can be destroyed if a temperature rises up to 25, gynecological  and genitourinary (bladder and prostate).This method is suitable for cancer cells treatment, which exists within or near the body while the applicator is inserted near the natural openings that may include gastrointestinal (esophagus and rectum), pulmonary (trachea and bronchus)Interstitial techniques are used to treat tumors deep within the body, such as brain tumors. When a single antenna fails to attain the necessary SAR uniformity over a tumor size for treatment zones that are substantial compared to penetration depth of the field at a particular frequency, arrays of antennas are used to treat such tumors.26. Cancer treatment for arms, legs, and some other body organs including the liver and lungs can be treated using regional perfusion techniques.Deep-seated tumors especially inside the abdomen and pelvis, require regional heating. Deep regional hyperthermia is performed through arrays comprising multiple antennas27:c Specific heat constant , t Exposed time in seconds.For the treatment of cancer spreading throughout the body parts, whole-body hyperthermia is considered a suitable option. All through the treatment of hyperthermia, the temperature rise is continuously monitored to ensure that desired temperature for destroying cancer tumors is achieved. For hyperthermia treatment, the successful treatment is determined by the rate at which the temperature in the body rises and the final value of temperature. SAR is calculated using the formula ast. SAR can also be calculated from the information of the Electrical field as28:However, Eq.  holds va29 where an electrically small antenna (ESA) was suggested comprising of Coplanar stripline (CPS) and capacitively loaded loops (CLL) to achieve quasi-isotropic radiation pattern. Antenna dimensions were taken as 20.6 mm Equation  is suita30. Antenna size was taken as 32 mm Antenna for hyperthermia applications require antenna performance in terms of Specific Absorption Rate(SAR) along with Effective Field Strength (EFS) and penetration depth (PD). A microwave lens applicator was recommends for the treatment of hyperthermia applications in Ref.31, suggested a sub 1 GHz antenna designed for hyperthermia. Several shapes were tested for antenna performance that, includes a rectangular patch, bow tie, truncated bow-tie, fishtail bow-tie and spiral. Antenna dimensions for the rectangular-shaped patches were 26.5 mm 32 for Hyperthermia applications at 2.45 GHz. Antenna dimensions were 29 Another work represented in Ref.33 to treat breast tumors placed around the breast to focus power inside the tower. Authors have used semi-ellipsoidal and conical-shaped phased array applicators having dimensions of 178 mm Phased array antennas are suggested in Ref.34, in which 18-element phased arrays operate at 434 MHz. Using Genetic Algorithm, 18, 15, 12 and 9 antennas were activated. The temperature of Another phased array applicator for breast cancer was suggested in Ref.35, discussed the design and fabrication of a compact, ultra-thin electromagnetic bandgap (EBG) backed antenna for the 24 GHz ISM band. The proposed antenna has a Koch fractal geometry-based bow-tie slot design, backed by a 5 Article36.A low profile hexagonal-shaped microstrip patch antenna was designed using FR-4 substrate, with dimensions of 124 mm 37. The applicator was made up of a double spiral antenna, an artificial magnetic conductor, and a frequency-selective surface. The applicator includes a frequency-selective surface at the top of the double spiral antenna to direct energy toward the tumor and ensure uniform heating. The research also explored the applicator’s ability to heat deeply-seated tumors, with the outcomes demonstrating a temperature profile of 44 A compact meta-material-based applicator specifically designed for hyperthermia cancer treatment was proposed in Ref.38, for Gastrointestinal track impedance response. Dimensions of dipole and loop are taken as 1.76 mm A meandered dipole and loop implantable antennas operating at 434 MHz, 1.4 GHz, and 2.4 GHz was put forward in Ref.39, multiple silicon layers are suggested to improve the SAR centralization suitable for hyperthermia applications. Antenna dimensions are taken as 1.93 mm In Ref.19, a patch antenna with circular rings was proposed with dimensions of 130 mm In Ref.40 operating between 400 and 800 MHz. The EFS of the antenna exceeds its dimensions (30 mm An open ridged-waveguide antenna (ORWA) was proposed to achieve ultrawideband in Ref.28 at 434 MHz. PD of 12 mm while EFS of 180 mmA microwave applicator for the treatment of gynecological cancers was suggested in Ref.30, lens applicator with dimensions of 32 The ISM band of 2.4 GHz is also frequently used for biomedical antenna design. In Ref.41, at 915 MHz and 2450 MHz. Results indicated that 915 MHz was suitable for stage 1, 2 and 3 cancer, while 2450 MHz was only able to perform well for early-stage cancer. Antenna dimensions at 915 MHz and 2450 MHz are taken as 87.66 mm For Breast cancer hyperthermia, a microstrip applicator designed on RO 4350  is proposed. Antenna operates at 434 MHz where miniaturization is achieved through a novel single lumped element scheme as compared to conventional multiple inductors and capacitors. A simplified design is proposed without any addition of Balun transformer or complex matching network. The EFS and penetration depth results exhibited that the proposed antenna is suitable for hyperthermia and implantable biomedical application. FDPA without lumped element produces a resonance notch at 2.65 GHz. The size reduction through lumped inductor has shifted the resonance notch at 434 MHz band, which is about 83% reduction in the size of the antenna.The manuscript is organized as follows. “This section comprises the geometry and optimized dimensions of FDPA. A parametric sweep is applied to achieve the optimized antenna performance.The proposed FDPA design comprises of folded dipole patch at the top layer of substrate material while two parallel strips at bottom layer of the substrate are excited through coaxial feed as illustrated in Fig. An inductor (lumped element) is added to the front layer of FDPA. Several positions are changed for placing the inductor in FDPA. However, there is no significant change noticed due to the variation in inductor position, hence the inductor is added at the centre of the left length of the patch. Moreover, the gap for this inductor is kept around 1 mm. An inductor value of 200 nH with 603 package is considered suitable for this design as it can be easily soldered to this small patch width of FDPA strip. Self-resonant frequency (SRF) of the inductor should be always greater than the resonance frequency of the antenna. As the final resonance occurs at 434 MHz therefore, the inductor package chosen comprises SRF greater than 434 MHz, which in this case is taken around 800 MHz with a package length 1.2 mm suitable for ESA. Antenna dimensions are summarized in Table The optimized return loss curve for the proposed FDPA is shown in Fig. Initially, coaxial feed is applied to the parallel strips at bottom of the substrate layer, while no lumped component is added to the FDPA. Here, the position of the feed matters. Antenna resonates at 2.65 GHz band with Applying parametric sweep to the back parallel strip length In the second step, four lumped elements are added in the design comprising of two inductors and two capacitors, to shift the resonance frequency to sub 1 GHz band, while the feed is provided to the parallel strips placed at the bottom layer of the substrate. The position of lumped elements on the top layer of the folded dipole patch antenna is illustrated in Fig. Figure To shift the impedance to sub 1 GHz, an inductor is introduced in the design of FDPA. The position of the inductor on the vertical length of the folded dipole does not create any difference in the shifting of the resonance frequency. Figure Figure Variation in The variations in the length of parallel strips . The appropriate value of the Lumped element (inductor) is chosen after various parametric sweeps. A bandwidth of 6 MHz (431–437 MHz) is achieved in this design.The fabricated antenna is presented in Fig. The radiation patterns for the proposed antenna in both E and H planes are illustrated in Fig. For measurements of SAR, PD and EFS, the proposed FDPA is placed near the three-layer phantom comprising skin, fat and muscle layer. Each layer has its own permittivity, conductivity and density as mentioned in Table Initially, the antenna is placed at a 3 mm distance from the three-layer phantom comprising of skin, fat, and muscles having thicknesses of 1 mm, 2 mm and 25 mm respectively. SAR measured is 3.59 W/kg at 0.5 W input power. EFS of 17 mm SAR for various orientations of antenna plane with respect to the phantom plane is presented in Fig. mentclasspt{minimaAs the antenna contains no ground plane, therefore maximum radiation direction of the antenna is not along broadside, instead, the maximum radiation is noticed at 130The shift in resonance frequency also depends on the distance of the antenna from the body (phantom). For the case of FDPA, a minor shift is observed between 0.42 and 0.43 GHz, as antenna distance is varied between 3 and 6 mm, as indicated in Fig. This work presents a miniaturized FDPA suitable for hyperthermia and implantable biomedical applications. The proposed antenna is electrically small, where impedance matching is achieved without any complex matching circuit or Balun transformer. Initially, the antenna resonates at 2.65 GHz while changing the position of the feed point and adding the lumped element to the folded dipole patch; it resonates in sub 1 GHz band at 434 MHz. Later the SAR, EFS and PD of the antenna in 3-layer human phantom is also calculated for a distance of 3 mm, 6 mm and 9 mm between the FDPA and the phantom. The proposed FDPA provides good performance for SAR, EFS and PD with the additional advantages of simple design and miniaturized size.


This article presents an antenna with compact and simple geometry and a low profile. Roger RT6002, with a 10 mm × 10 mm dimension, is utilized to engineer this work, offering a wideband and high gain. The antenna structure contains a patch of circular-shaped stubs and a circular stub and slot. These insertions are performed to improve the impedance bandwidth of the antenna. The antenna is investigated, and the results are analyzed in the commercially accessible electromagnetic (EM) software tool High Frequency Structure Simulator (HFSS). Afterwards, a two-port multiple–input–multiple–output (MIMO) antenna is engineered by orthogonalizing the second element to the first element. The antenna offers good value for mutual coupling of less than −20 dB. The decoupling structure or parasitic patch is placed between two MIMO elements for more refined mutual coupling of the proposed MIMO antenna. The resultant antenna offers mutual coupling of less than −32 dB. Moreover, other MIMO parameters like envelop correlation coefficient (ECC), mean effective gain (MEG), diversity gain (DG), and channel capacity loss (CCL) are also studied to recommend antennas for future applications. The hardware model is fabricated and tested to validate the results, which resembles software-generated results. Moreover, the comparison of outcomes and other important parameters is performed using published work. The outcome of this proposed work is performed using already published work. The outcomes and comparison make the presented design the best option for future 5G devices. The swift advancement in communication systems has been recognized since the last decade. The transformation of third generation 3G) to fourth generation 4G or LTE), and then 4G to fifth generation 5G), and now the current deployment of future sixth generation (6G) all demonstrate many changes in communication models and systems ,2. TheseG to fourG or LTE,G, and noBeing an important and essential component of the communication model, antenna design requirements were revised due to changes in communication devices or models . To meetResearchers and academics have suggested a variety of antennas for this purpose that operate at high gain, wideband, and compact sizes . Multi-iIn many MIMO antennas, the required value of mutual coupling is not met, for which various techniques are used to obtain the mutual coupling below 20 dB. Researchers have adopted many techniques to improve the isolation between elements in MIMO antenna systems. These techniques include the loading of FSS (Frequency Selective Surface) sheets ,16, absoIn the literature, the metamaterials, the defective ground structure (DGS), and the decoupling structures are widely used to upgrade the outcomes of the MIMO antenna in terms of mutual coupling, bandwidth, and gain. To enhance mutual coupling, the antenna is loaded with metastructures, either of one particular layer (same layer or substrate) or a distinct one (load another substrate) . The antIn ,22, a hiThe most efficient technique employed to lessen mutual coupling between MIMO antenna elements is loading decoupling structures between MIMO elements, or parasitic patches/element . In the A DRA antenna loaded with a parasitic patch is given in , which oCompact size and simple design structure;Wideband and high gain;Simple and new decoupling structure;Low mutual coupling;Low value of ECC and acceptable value of other MIMO parameters.A thorough study of the literature reveals that there is still a need for research to develop antennas with small sizes, straightforward geometries, and low profiles, as well as strong gains, wideband, and low mutual coupling. Due to this fact, a unique and straightforward decoupling structure for the antenna is described in this study in order to eliminate mutual coupling. The presented design of an antenna has the following advantages over other work presented in the literature:The rest of the paper is divided into sections. In next section, the single element of the antenna is discussed along with its results. The fabricated antenna prototype and comparison between measured and simulated results are given in X × AY × t = 10 mm × 10 mm × 1.52 mm. The antenna is excited using coaxial cables with 50 Ω impedance matching. The coaxial wire connects to the antenna after passing through the substrates. Using a coaxial cable has the advantages of low cost, being easy to wire and expand, and being supportive of high-bandwidth signals [1 = 3 mm; R2 = 2.7 mm; R3 = 2.5 mm; FX = 3.5 mm; FY = 0.5 mm; t1 = 1.52 mm. The antenna is analyzed, and its parameters are studied using an electromagnetic (EM) software tool, High Frequency Structure Simulator (HFSSv9). The geometrical and structural layout of the presented dual-port MIMO antenna is given in  signals . The res1 = 2 mm is obtained using the circular patch antenna equation given in [In the initial design stage, a circular patch antenna with a coaxial feedline is engineered for 28 GHz applications. The radius of the circular patch Rgiven in . The ant2 = 2.5 mm. After this step, the antenna starts resonating at 28.2 GHz, with an impedance bandwidth of 27.1–29.2 GHz. The value of return loss also improved after this step and approached 27 dB.To upgrade the bandwidth and refine the return loss of the antenna, another circular stub is loaded onto the antenna, which has a radius of RFor further improvement in results, a circular-shaped slot with a radius of R = 2.5 mm is etched from the antenna. After this step, the final structure of the antenna is obtained, and it starts operating at a broadband of 25.25–29.85 GHz. At this stage, the antenna offers a return loss of –45 dB.In this manuscript, a broadband antenna with an efficient value of return loss is proposed for future 5G devices. Desired and beneficial outcomes are achieved after going through three design steps. The design process of the proposed antenna is divided into the three stages shown below:11 plot is given in The antenna design evolution along with its impact in terms of the SThe performance evaluation of an antenna is analyzed by studying its S-parameter curve and gain versus frequency curve. It is noticed from The radiating property of the presented broadband design is examined by studying the radiation patterns of proposed antenna. The radiation pattern of a single element of this work is provided in X × MY = 10 mm × 26 mm. Both of the elements are fed by a coaxial cable and are shown in A = 10 mm, PB = 1 mm, PC = 0.5 mm, PD = 0.5 mm, and PE = 5 mm. To lessen the mutual coupling between MIMO parts, the decoupling element is mounted onto the antenna.For the further validation of outcomes, an antenna prototype is constructed and tested. A snap of the antenna prototype is given in The antenna is transformed to a MIMO configuration by placing another antenna element orthogonal to the reference antenna. The two components of the MIMO antenna are separated by an 8 mm gap. The antenna has the same optimal characteristics as the single antenna element covered in In Radiation pattern is the key parameter used to study the radiating properties of a two-port antenna. The antenna’s radiation pattern is shown in For more validation of the recommended MIMO antenna design for millimeter applications, some important MIMO parameters are studied and measured. Envelope correlation coefficient (ECC), mean effective gain (MEG), diversity gain (DG), and channel capacity loss (CCL) are some of these parameters.ECC shows the performance of independent antennas in a MIMO system. Ideally, the value of ECC should be equal to zero or approximately equal to zero and can be calculated using the formula given below .(1)ρeijFor this design, the value of ECC is less than 0.0001 through an operational bandwidth of 25.25–29.85 GHz, as given in Rψ refers to the below matrix for receiving antenna correlation and its mathematical relation is given below.Another important MIMO parameter, which is studied in this paper, is channel capacity loss (CCL). The study of correlation losses in MIMO systems is known as CCL. Less than 0.5 bits/Hz/s should be the CCL value, as estimated by the equations provided below .(2)CCL=The CCL of the suggested MIMO antenna is shown in Diversity gain (DG) is one of the most important parameters of MIMO antenna systems, which measures the losses experienced in the diversity scheme, and is the most crucial metric in MIMO systems. The permissible diversity increase value can be calculated by Equation (4) and its maximum value is 10 dB; however, in real-world situations, a value of roughly 10 dB is acceptable . The matAccording to Mean effective gain (MEG) is a key metric of the MIMO system, which shows the received power in the fading area. The acceptable range of MEG is less than –3 dB. The proposed work offers MEG less than –5 dB, which is under the acceptable range.From the above discussion, it is clear that the proposed MIMO antenna offers an acceptable range of MIMO parameters. A comparison between simulated and measured MIMO parameters is also provided. The value offered and the comparison show that the proposed antenna is a good candidate for future devices, which operate at high data rate.The results in the form of bandwidth, gain, isolation, and ECC of the proposed antenna are compared with published work to prove the antenna’s superiority. In the literature table given below, the antenna given in ,30,31 haTwo-port MIMO antennas for 28 GHz are presented in this article. Initially, a single element was designed after following three design steps. Afterwards, the two-port MIMO antennas were generated, giving a mutual coupling less than −20 dB. The decoupling patch was loaded between two antenna elements to overcome mutual coupling. After loading the decoupling structure, the antenna offered mutual coupling less than –32 dB. The MIMO antenna presented in this article has a compact size of 25 mm × 10 mm, with a thickness of 1.52 mm. The antenna operates at a wideband of 25.25–29.85 GHz with a peak gain of 10.8 dBi. The antenna offers a good value of ECC, around 0.001, and DG around 9.99 dB. Other MIMO parameters are also analyzed, which are in the acceptable range. The software study of the design and analysis of the antenna was carried out using the EM software tool HFSS (High Frequency Structure Simulator). The hardware prototype was fabricated to validate the simulated outcomes. Moreover, a comparison is provided in the form of table to compare the antenna results with published works. The outcomes and table of comparison make the proposed antenna a potential applicant for future 5G millimeter-wave applications.


This paper presents a printed multiple-input multiple-output (MIMO) antenna with the advantages of compact size, good MIMO diversity performance and simple geometry for fifth-generation (5G) millimeter-wave (mm-Wave) applications. The antenna offers a novel Ultra-Wide Band (UWB) operation from 25 to 50 GHz, using a Defective Ground Structure (DGS) technology. Firstly, its compact size makes it suitable for integrating different telecommunication devices for various applications, with a prototype fabricated having a total size of 33 mm × 33 mm × 0.233 mm. Second, the mutual coupling between the individual elements severely impacts the diversity properties of the MIMO antenna system. An effective technique of orthogonally positioning the antenna elements to each other increased their isolation; thus, the MIMO system provides the best diversity performance. The performance of the proposed MIMO antenna was investigated in terms of S-parameters and MIMO diversity parameters to ensure its suitability for future 5G mm-Wave applications. Finally, the proposed work was verified by measurements and exhibited a good match between simulated and measured results. It achieves UWB, high isolation, low mutual coupling, and good MIMO diversity performance, making it a good candidate and seamlessly housed in 5G mm-Wave applications. There is no doubt that this era is one of wireless communication technology that provides its contributed to the development of very high data rates, higher capacity, and reliability in communication applications used daily by a wide range of people ,2. RecenConcerning the separation between the antenna elements, the mutual coupling is inversely proportional to it. However, this distance should be chosen to be at a minimum between λ/4 and λ/2 to have a mutual coupling of less than −15 dB and therefore have an efficient working MIMO system. Additionally, a larger MIMO system size results from the increased separation between antenna units. As a result, while designing MIMO antennas, trade-offs between the total size of MIMO systems and the minimizing of mutual coupling should be taken into account . Many researchers aim to enhance the performance of different parameters such as bandwidth, compact size, gain, efficiency, mutual coupling and diversity properties in the Fifth Generation (5G) Millimeter-Wave (mm-Wave) antenna design. So, they use several enhancement techniques to obtain the best performance, like substrate choice, corrugation, multi-element, dielectric lens and mutual coupling reduction techniques ,9. FirstA variety of approaches have been used to reduce mutual coupling between MIMO antenna elements. Neutralization techniques, simultaneous matching of orthogonal feeding or elements, pattern diversity, and significant polarisation are all included ,12,13. IOn the other hand, Diversity properties of MIMO systems like envelope correlation coefficient (ECC), Channel capacity loss (CCL), Diversity Gain (DG), Multiplexing Efficiency (ME) and Total Active Reflection Coefficient (TARC) are strongly affected by the mutual coupling between antenna elements of MIMO system ,28,29,30Although significant work and research were made in the last half-decade to introduce reliable MIMO antennas with acceptable diversity parameters, In this work, the DGS and antenna element orientation are developed to achieve acceptable isolation along the operating band. In this study, a 4-port MIMO antenna system covering the frequency range of 25 to 50 GHz is developed, constructed, and tested. The ground plane’s carved slots between the antenna elements and the orthogonal orientation of the antenna elements form the design’s foundation. A DGS is then used to deploy for mutual coupling reduction and gain enhancement after a reference antenna is tuned for bandwidth enhancement. The work is organised as follows; Since the antenna is the pivot element of MIMO antenna systems, many researchers present different antennas adapted for this technology. Concerning mm-Wave applications, printed antennas are the best candidates for this technology because of their low cost, low profile, compact size, and good choice to achieve a functional element with a proper balance of performance and manufacturing complexity for millimeter wave applications.An overview of the sequential design evolution is shown in ned from :(1)W=c2fThe rectangular patch antenna is improved for a larger band in the second step by etching the ground (DGS) and adding vertically and horizontally oriented circular slots to the three sides, as shown in The long microstrip feed in The fabrication process includes many steps. Based on the available dielectric substrate, we select the appropriate one suitable for the required frequency band. The antenna is modelled on CST microwave studio simulator. The optimization process is developed. Then, the Gerber file and “.Sab” file are exported to the manufacturer. At this step the duty of the academic researchers is finished. On the fabrication side the mask of design is developed. The copper is etched from the top and bottom surfaces using chemical process.When presenting a MIMO antenna along with other fundamental characteristics, Diversity parameters analysis is a must in addition to the fundamental antenna performance evaluations and parameters, such as bandwidth, resonance frequency, radiation patterns, gain, and efficiency. These characteristics are necessary for multi-antenna devices but not for single-antenna elements. The parameters used for this include the TARC, CCL, DG, and ECC The next subsections go over these variables in relation to the suggested antenna design.The antenna is fabricated as shown in The level of correlation between various antenna components in a MIMO configuration is measured by ECC. Lower ECC results in less interdependence between the components and, as a result, better MIMO diversity performance. Equations (6) and (7), respectively, show how to compute the ECC from both the far field radiation pattern and the scattering parameters ,35.(6) e useful . HoweverDG value is obtained from ECC, and it can be easily calculated using the following equation:It is clear that ECC and DG are inversely proportional to one another. A high DG value therefore indicates superior performance. The DG may be determined using both the far field radiation pattern and the scattering parameters, much like the ECC can . Better The CCL parameter, which specifies the highest achievable limit of the information transmission rate, can be used to describe the channel capacity of MIMO systems. It calculates the ideal information transfer rate, in other words. An Equations from  can be uME is defined as the ratio between the power of the real antenna and that of the ideal one. The maximum ME has been calculated as :(13)ηmaxThe ratio between the square roots of the total power reflected and the total power incident, is known as the total active reflection coefficient (TARC). It may be estimated using the S-parameters and depicts the random signal combinations and mutual couplings between the ports. TARC is the sole MIMO parameter that takes into account the unpredictable phases of incoming signals, which can have a significant impact on MIMO array behavior in certain circumstances. TARC of 4-port MIMO antenna can be calculated as ,40:(14)TThe value of TARC for a MIMO communication system shouldn’t exceed 0 dB. In The proposed MIMO antenna system has been designed to provide high isolation and UWB capability, with a perfect alignment orientation between the four elements and etching the ground using DGS. The proposed antenna system has achieved excellent performance parameters, such as low mutual coupling of less than −10 dB, low ECC of less than 0.005, low CCL of 0.25 bits/s/Hz, high DG of 9.999 dBi and low TARC of less than -10 dB, which shows a superior diversity performance. In addition, the wide operational bandwidth of 25 GHz, from 25 to 50 GHz, enables the antenna to be used in multiple applications. The compact design and low-cost design features make it suitable for mm-wave 5G applications and can be easily integrated into telecommunication devices. Furthermore, the suggested MIMO antenna system offers great performance and portability, making it an ideal choice for a wide range of 5G mm-wave applications.


Furthermore, a radiation efficiency of 94.2% and 89.7% is realized at 1.7 and 2.5 GHz, respectively. The proposed antenna attains a measured average gain of 5.2 dBi and 6.1 dBi at the L and S band, respectively.In this paper, a printed monopole antenna with high-gain and dual-band characteristics for applications in wireless local area networks and the internet of things sensor networks is presented. The proposed antenna consists of a rectangular patch with multiple matching stubs surrounded to improve the impedance bandwidth of the antenna. The antenna incorporates a cross-plate structure which is seated at the base of the monopole antenna. The cross-plate consist of metallic plates aligned perpendicularly which enhances the radiations from the edges of the planar monopole to maintain uniform omnidirectional radiation patterns within the antenna’s operating band. Furthermore, a layer of frequency selective surface (FSS) unit cells and a top-hat structure is added to the antenna design. The FSS layer consist of three unit cells printed at the back side of the antenna. The top-hat structure is placed on top of the monopole antenna and comprises of three planar metallic structures arranged in a hat-like configuration. The coupling of both the FSS layer and the top-hat structure presents a large aperture to increase the directivity of the monopole antenna. Thus, the proposed antenna structure realizes a high gain without compromising the omnidirectional radiation patterns within the antenna’s operating band. A prototype of the proposed antenna is fabricated where good agreement is achieved between the measured and full-wave simulation results. The antenna achieves an impedance bandwidth |S Microstrip antennas with compact sizes, wide bandwidth and high-gain performance features are desired for most wireless communication systems3.The advancement of modern wireless systems continually places stringent requirements especially on the performance of antennas. Moreover, the rapid development of mobile communication systems and the wireless local area network (WLAN) also require antennas with high performance, omnidirectional radiation patterns and dual-band characteristics7. Conventional printed monopole antennas however suffer from low gain and stable omnidirectional radiation patterns which limit their operation range8.Printed monopole antennas have received a lot of attention due to their omnidirectional radiation pattern characteristics in addition to their low cost and low profile features which makes them suitable candidates for many wireless applications including the internet of things (IoT) and WLAN10. While these methods are effective gain enhancement techniques, they often result in antennas with large structures.Traditional gain enhancement methods which involve an extension of the length of the monopole patch radiator or increasing the ground plane size have been proposed in a number of studies15. For example, an HIS structure composed of periodic square patches is implemented below the ground plane of a fork-shaped patch antenna to increase the antenna gain11. In Ref12, a wearable antenna which consists of a monopole antenna incorporated on an AMC structure for high-gain properties is presented. Also, a flexible and frequency reconfigurable monopole antenna with an FSS structure is presented for IoT applications13. The gain of the antenna is enhanced by placing an FSS structure beneath the antenna. In Ref14, a monopole directional antenna with a bioinspired elliptical leaf configuration is presented. The omnidirectional radiation pattern is converted to a directional pattern by placing a reflector near the ground plane. Again, in Ref15, a double split ring metasurface reflector is employed to enhance the gain of a monopole antenna. Although a significant gain enhancement is realized, large and complex structures are realized. Moreover, these gain enhancement methods result in bulky configurations since the distance between the reflective surfaces must correspond to about one-half of the wavelength.A popular gain enhancement method used nowadays involves the addition of reflective layers very close to the antenna’s radiating structure. These structures are often composed of metamaterial or periodic structures such as artificial magnetic conductors (AMC), high impedance surface (HIS), and frequency selective surfaces (FSSs) arranged in a particular geometrical manner to provide in-phase reflection to the incident signal which improves the antenna gain17. A high-gain circularly polarized antenna, which utilizes a PRS structure placed above the microstrip patch antenna is presented in16. Also, a high-gain cavity resonator antenna using a metamaterial superstrate is proposed in17. Although a high-gain is attained with this method, large and complex antenna structures are realized. Moreover, the antenna gain enhancement is dominant only in a particular direction, thereby the omnidirectional pattern characteristics of the monopole antenna is distorted.Furthermore, the method of using superstrate layers such as partially reflecting surface (PRS) and metasurface structures to enhance the directivity of microstrip patch antennas have been presented in a number of studies23. Metamaterial structures placed on or around the radiating patch of the antenna have been explored for high-gain while maintaining the compactness of the antenna20. Moreover in21, a circular ring-shaped monopole patch antenna on a hexagonal ground plane has been presented for high-gain characteristics. Although these structures attained high-gain with compact sizes, the omnidirectional radiation patterns are usually distorted. In Ref22, a compact monopole antennas with a sleeve ground plane is presented. The sleeve coupled with the ground plane form a quarter-wavelength cavity which enhances the gain. Although a stable omnidirectional radiation patterns is realized, low gain values were recorded.To attain compact high-gain monopole antennas, several antenna structures have been proposed in a number of worksA closer look at the various gain enhancement methods proposed in prior works usually involve the addition of structures that results in high-gain monopole antennas with one-sided high directive beam patterns or distorted omnidirectional radiation patterns.In this paper, we present a novel method of realizing a printed monopole antenna with high-gain characteristics using a layer of FSS and a top-hat structure. The proposed antenna realizes high-gain with stable omnidirectional radiation patterns within the operating band of the antenna. Furthermore, the proposed monopole antenna realizes dual-band characteristics which makes it suitable for WLAN and various wireless applications including the IoT sensor networks.h) of 0.508 mm and loss tangent (tan δ) of 0.0009. The top-hat structure is composed of copper metal plate arranged in a hat-like manner and is seated on the TLY-5 Taconic substrate, at the top of the monopole antenna. At the bottom part of the Taconic substrate, a metallic plate is inserted perpendicular to the printed partial ground plane at the backside of the antenna to form a cross-like printed ground plane structure.The configuration of the proposed monopole antenna is given in Fig. In the antenna design, the patch radiator is surrounded by stubs placed at optimized distances to improve the impedance bandwidth of the antenna. The rectangular patch is fed by a feedline matched to a 50 Ω-SMA connecter via an impedance transformer for measurement. The antenna is situated on a circular base with radius 30 mm, which also serves as a support to the antenna structure. The detailed dimensions of the antenna structure consisting of the patch, FSS unit cells and stubs are illustrated in Fig. g (where λg corresponds to the guided wavelength). For applications in the maritime environment, the base station is usually located several meters above the antenna system embedded on the boats, thereby requiring antenna structures with tilted beam patterns. The current distributions for a dipole or monopole antenna of length 1.5λ and 0.75λ, respectively results in the tilting of the maximum beam in the upward direction. Therefore in this work, the length of the monopole patch radiator is designed at 3/4 λg (a derivative of 1/4 λg) in order to generate tilted radiation pattern beams. The conventional monopole antenna exhibits an impedance bandwidth of |S11| < ‒10 dB at 1.35 to 1.6 GHz and 2.1 to 2.4 GHz. Moreover, a low gain below 3 dBi is realized at the lower frequency band of the antenna.The conventional monopole antenna is composed of the radiating patch, feedline and the circular base. The radiating patch for the conventional monopole antenna is usually designed at a quarter-wavelength i.e., 1/4 λl4 and width w3.To improve the impedance matching conditions of the conventional monopole antenna, open-circuited rectangular stubs are added to the sides of the rectangular patch radiator as illustrated in Fig. 11) and voltage standing wave ration (VSWR) are presented and compared to the conventional one in Fig. 11| < ‒10 dB and VSWR ≤ 2 at the lower frequency band of 1.32 to 2.3 GHz, and an upper frequency band of 2.6 to 2.9 GHz is achieved with the addition of stubs.The results of the simulated reflection coefficient (Sxoy-plane) does not have a uniform omnidirectional pattern. This is because the radiation pattern is more directive in the yoz-plane compared to the xoz-plane. Furthermore, the omnidirectional radiation patterns of the conventional monopole antenna distorts with frequency variation within the antenna’s operating band. One way to have uniform omnidirectional radiation patterns is to reduce the width of the patch radiator. However, this approach reduces the bandwidth characteristics of the antenna.One of the drawbacks of the conventional printed monopole antenna is that the azimuth radiation pattern .To address this challenge, a cross-plate is incorporated into the conventional monopole antenna as shown in Fig. wcross which plays a critical role in attaining a flat omnidirectional radiation pattern is varied. As shown in Fig. wcross. It can also be observed that the best azimuth radiation pattern flatness is achieved when wcross = 44 mm. However, the variation of wcross seems to have no significant impact on the impedance bandwidth of the monopole antenna as depicted in the S11 and VSWR results in Fig. To further study the physical significance of the cross-plate, the width of the cross-plate, A, while the conventional monopole with both stubs and cross-plate added is labelled as Structure B. From the azimuth results, it can be observed that stable radiation patterns are attained when the cross-plate is added compared to the conventional monopole antenna. Furthermore, the 3D-peak gain results shown in Fig. The azimuth radiation pattern results are given in Fig. To enhance the gain of the conventional monopole antenna especially with regards to the lower frequency band , a top-hat structure is added at the end of the monopole antenna. The configuration of the top-hat structure is given in Fig. The motivation behind using the metallic top-hat is to present a large radiating aperture to the monopole antenna to enhance the directivity, without increasing the overall size of the antenna. However after the incorporation of the top-hat structure, the realized gain of the monopole antenna deteriorates, although a high directivity is attained. This is mainly attributed to the poor matching conditions of the antenna after the addition of the top-hat.24. The unique frequency-shifting ability of the FSS is employed to improve the matching conditions of the antenna due to its simplicity and ease of fabrication. Moreover, the FSS structures have additional advantages of resonant properties that could be utilized to improve the impedance bandwidth characteristics of the antenna. The FSS unit cell can behave as transmissive and reflective surface based on the dimensions. Therefore, each unit cell is designed with optimized dimensions of 34 mm × 34 mm (0.3λg × 0.3λg) to achieve the desired characteristics of the target band of the monopole antenna.In order to reap the full benefits of the top-hat structure while attaining good matching conditions of the antenna, a layer of frequency selective surface (FSS) consisting of three square-loop metallic unit cells are placed at the backside of the monopole radiating patch. FSS structures are frequency dependent surfaces composed of periodic structures that can manipulate the characteristics of the electromagnetic waves passing through it11| and VSWR results are given in Fig. 11| < ‒10 dB and VSWR ≤ 2 at 1.35–2.1 GHz and 2.4–2.75 GHz.The top-hat structure coupled with the FSS unit cell layer results in a peak gain of about 5.2 dBi at the target band as depicted in Fig. LHAT, SHAT and WHAT, respectively), a detailed parametric study in terms of the reflection coefficient (S11), VSWR and realized gain is given in Fig. LHAT) as shown in Fig. LHAT = 40 mm.To further study the effect of the key design parameters of the top-hat structure i.e., the length, width and spacing  seems to increase the gain of the antenna at the target frequency band as shown in Fig. WHAT variation as the impedance bandwidth performance increases with decreasing WHAT. In this case, the best performance considering both gain and impedance is realized when WHAT = 30 mm.Moreover, increasing the width of the top-hat (SHAT) is varied to determine its impact on the gain and impedance bandwidth performance as shown in Fig. SHAT as in the case of WHAT that, a trade-off occurs between the gain and impedance bandwidth performance. From the results in Fig. SHAT = 30 mm.Furthermore, the spacing between the vertical metallic plates of top-hat  and at 2.4 to 2.85 GHz (BW = 17.14%).The simulated and measured |SIn addition, the simulated and measured gain results of the proposed antenna are compared in Fig. xoy-plane), elevation (xoz-plane) and elevation (yoz-plane) radiation patterns, respectively at 1.7, 1.8, 1.9 and 2 GHz. It can be observed from the azimuth radiation patterns that the proposed antenna maintains a stable and uniform omnidirectional radiation patterns within the antenna’s operating band. Furthermore, a 3 dB-beamwidth of 46.4°, 49.0°, 41.7° and 36.0° at 1.7, 1.8, 1.9 and 2.5 GHz, respectively is observed for the elevation (xoz-plane) shown in Figs. yoz-plane) radiation patterns of the proposed antenna. Both the measured and simulated patterns agree well which shows that the proposed antenna attains stable and uniform omnidirectional radiation patterns within its operating band.Moreover, Figs. 11, however distorted omnidirectional patterns are observed for Ref11. In addition, the proposed antenna realizes a good impedance bandwidth performance compared to the reference antennas. Also, the proposed antenna achieves a good radiation efficiency comparatively. The high-gain characteristics with the added advantages of stable omnidirectional radiation patterns highlights the proposed antenna as suitable for application in wireless communications and WLAN sensor networks.The performance of the proposed high-gain monopole antenna is compared with existing monopole antenna structures in Table In this paper, a novel method of enhancing the gain of a printed monopole antenna while maintaining the omnidirectional radiation pattern characteristics throughout its operating frequency band has been presented. The presented antenna structure is composed of a rectangular patch surrounded by multiple stubs for impedance bandwidth enhancement. To attain good omnidirectional patterns within the antenna’s operating band, a cross-plate structure is incorporated into the base of the antenna to enhance the edge radiations of the monopole antenna thereby preventing shrinking of the omnidirectional radiation patterns. Furthermore, a top-hat and a layer of FSS unit cells, printed at the backside of the antenna is incorporated, which increases the effective aperture of the antenna and enhances the antenna gain. The proposed antenna realizes dual-band characteristics in the L and S band at 1.6 to 2.1 GHz, and 2.4 to 2.85 GHz, respectively. A peak gain of 5.2 dBi and 6.1 dBi is realized for the L and S band, with an average radiation intensity of 94% and 89%, respectively. The proposed monopole antenna has a high-gain with the added benefits of stable omnidirectional radiation patterns which makes it suitable for applications in wireless communications and IoT sensor newtorks, and satisfies the stringent requirement of omnidirectional patterns needed for WLAN applications.


The proposed low-cost antenna is easily implemented in a typical university lab-based environment. The total bandwidth for the three modes is close to 1 GHz, while the voltage standing wave ratio (VSWR) of the fabricated version of the antenna does not exceed 1.02, and the return loss is well below −40 dB for the three primary resonant frequencies.This paper presents a compact multifrequency reconfigurable patch antenna in terms of design and fabrication for operating in the S and C bands of the RF spectrum, which are overwhelmed by wireless applications. Reconfiguration is achieved by using a single PIN diode on the ground plane. By varying the voltage applied to the diode, three modes can emerge, exhibiting main resonant frequencies at 2.07, 4.63, and 6.22 GHz. Resonance switching requires a voltage of less than 0.9 V. The antenna fabricated on an FR-4 substrate, with a volume of 70 × 60 × 1.5 mm The rapid growth of wireless communications has recently led to the demand for platforms supporting several communication standards. Such platforms can benefit highly from reconfigurable antennas, which can function in multiple bands, radiation patterns, or polarizations. Reconfigurable antennas are currently used in cognitive radio systems, satellite communications, MIMO systems, biomedical applications, as well as military and industrial applications ,2. The pReconfigurable antennas can change their resonant frequencies, radiation patterns, polarization, or any combination thereof. Reconfiguration is achieved usually by one of the following methods: electrical switching, optical switching, mechanical switching, and smart materials.Reconfiguration via electrical switching is the most common method used. It involves PIN (positive–intrinsic–negative) diodes, varactor diodes, or RF MEMS . Optical switching concerns using photoconductive switches, while mechanical switching involves the physical movement of radiating parts ,6. FinalMany researchers have presented simulation results of reconfigurable antennas of various geometries . In someMost of the available literature concerning antenna reconfiguration characteristics via PIN diodes has so far concentrated on two or more PIN diodes ,24,25,26The presented antenna was designed for integration into a high-precision signal power measurement setup, akin to the one detailed in . This seThis paper is organized as follows. The structural geometry of the antenna is illustrated in The antenna simulation was conducted using CST Studio Suite 2019 with open-space boundary conditions. To provide excitation, a waveguide port was positioned at the edge of the microstrip feed line, with a port extension coefficient of 7.2. For the substrate material, lossy FR-4 was used, complemented by parts of annealed copper. The model of the PIN diode used is Skyworks SMP1302 in an SC-79 package. According to its datasheet, the diode can be used in applications from 10 MHz to beyond 10 GHz. In the simulation, the diode had to be represented by its equivalent circuit, because the simulation tool does not support nonlinear elements. The diode can be represented by an RL circuit when it is forward-biased (ON state) or by an RLC circuit when reverse-biased (OFF State), as shown in In the simulation, the resistance (R) and the capacitance (C) values were selected from the diode’s datasheet graphs, considering the voltage on the diode to be between 0 and 900 mV. L is always selected as 0.7 nH, as defined for the SC-79 diode package. Simulations encompassed the entire 0 to 900 mV range, corresponding to values of R ranging from 1 to 1000 Ω under forward-bias conditions. Notably, three distinct cases stood out, distinguished by the conspicuously low S11 parameter values observed at their primary resonant frequencies. These cases were denoted as modes 1, 2, and 3. Mode 1 corresponds to 0 voltage on the diode, with values C = 0.3 pF and R = 5 kΩ. Modes 2 and 3 correspond to a forward biasing of the diode with no C and R = 160 Ω or R = 4 Ω, respectively. The simulated S11 parameter and VSWR results for these three modes are illustrated in x-axis extends towards the front direction of the antenna. The azimuth angle will be referred to as φ, while the elevation angle as θ. Both are 0° at the antenna’s front while 180° at the back.As indicated by According to the 2.04 GHz radiation pattern, the maximum gain of the antenna is, in this case, 0.82 dBi, observed at angles θ = 355° and φ = 180°. Furthermore, in front of the antenna, for θ = φ = 0°, the gain is −0.78 dBi. Similarly, the 4.44 GHz pattern shows a maximum gain of −0.31 dBi, at angles θ = 310° and φ = 135°. In front of the antenna, at θ = φ = 0°, the gain is −15.74 dBi. However, the gain can be increased to −3.78 dBi with the antenna’s rotation to an angle of θ = 34° on the elevation plane. Finally, the 6.42 GHz radiation pattern shows a maximum gain of 5.68 dBi, at angles of θ = 20° and φ = 0° while at angles of θ = φ = 0°, it equals 4.13 dBi.As derived from the simulation, the antenna performs best when operating on mode 3, with a main resonance at 6.42 GHz. It demonstrates the maximum gain, and lowest return loss and VSWR.Finally, Last but not least, the simulations demonstrate a good radiation efficiency for the antenna across its three primary resonant frequencies. At 2.04 GHz, the radiation efficiency stands at an exceptional 74%, while at 4.44 GHz and 6.42 GHz, it remains notably high at 59% and 54%, respectively. These findings are consistent with typical performance expectations, signifying the absence of significant losses on the antenna.The simulation results for the three resonant frequencies, where the antenna seems to function optimally, are summarized in The front and the back side of the fabricated multi-frequency reconfigurable patch antenna is shown in A Rohde & Schwarz ZVH8 Handheld VNA was used to experimentally validate the performance of the multifrequency reconfigurable patch antenna through S11 and VSWR. It is also essential to mention that, in our setup, we utilized a high-quality DAC as the voltage source for biasing the diode. The DAC’s output has a low impedance of 1 Ohm, ensuring a stable DC bias voltage without significant AC components. Additionally, it provides exceptional isolation from AC signals, about 100 dB. Consequently, while the inclusion of chokes is essential in many cases, it was deemed non-critical for our application.The measurements of the antenna radiation pattern took place in a free-space lab environment in the university with dimensions of 5 m width by 10 m length. The free-space lab measurement plan appears in The S11 and VSWR results are depicted in Regarding the truncation of the VSWR plots , it is iNext, the radiation patterns for these three modes at 2.07, 4.63, and 6.22 GHz, on both the azimuth and the elevation planes, are illustrated in While the cross-polarization pattern can offer valuable insights into how the antenna interacts with signals of different polarizations, it is excluded  for the The experimental results indicate that the antenna operates almost equally well in all three bands of 2, 4, and 6 GHz, at the resonant frequencies of 2.07, 4.63, and 6.22 GHz. However, at 6.22 GHz, a much higher gain can be attained. A summary of the results, derived from the network analyzer measurements and the radiation patterns, is shown in As compared to the simulation results, some slight deviations are observed. In the simulation, the three main resonances were located at 2.04, 4.44, and 6.42 GHz, while, on the fabricated antenna, they were found at 2.07, 4.63, and 6.22 GHz. The deviation between the three cases is 30, 190, and 200 MHz, respectively, or 1.4%, 4.2%, and 3.1%.Also, the experimental results show a better functionality of the antenna at 4.63 GHz, than the one suggested by the simulation at 4.44 GHz. Furthermore, the radiation patterns of the fabricated antenna show better gains than those seen in the simulation. For the 2, 4, and 6 GHz bands, the maximum simulation gain was 0.58, 1.75, and 5.64 dBi, respectively, while the corresponding measured gains were found as 5.48, 3.97, and 12.79 dBi, which are 4.9, 5.72, and 7.15 dB higher than those suggested by the simulation. The deviations between the simulation and the experimental results are overall insignificant, except for the simulated and measured gain. These deviations were expected since many factors were not accounted for in the simulation. Such factors are the effects of solder used for the SMA connector, the PIN diode, and the effects of the pins used for applying voltage on the diode. Also, the antenna was tinned to avoid oxidation, which could also have some effect on its functionality. But more importantly, the deviations could be due to the R and C values selected for the diode’s equivalent circuit in the simulation. Both these parameters, but mostly R, are frequency-dependent. Because the simulation tool does not support nonlinear elements in the simulations, they were both considered constant over the whole simulated frequency range, from 0 to 8 GHz. Furthermore, the graphs in the diode’s datasheet describe these values for frequencies only up to 2 GHz. Thus, the R and C behavior for higher frequencies was considered to be similar, which might not be the case. This probably is the reason that the deviation appears to be minimal at the 2 GHz band. Finally, the deviations in the radiation patterns, and primarily in the gain, might be due to reflections occurring in the vicinity of the antenna, or due to the prime resonances of the fabricated antenna not being identical to those indicated by the simulation.The compact patch re-configurable antenna is controlled by a single PIN diode, in contrast to several other works that require multiple diodes. The antenna has three modes of operation with a total bandwidth of 974 MHz over prevailing tuning in the three frequencies. Furthermore, it has a low cost, volume, and weight and can be easily implemented even in a lab-based environment. The antenna also achieved quasi-omnidirectional radiation patterns that make it ideal for wireless applications where omnidirectional coverage is usually required.As we have already mentioned, there are several techniques to design and implement a reconfigurable antenna. The comparative study is based on antennas being similar to the one presented in this work. They are all printed on an FR-4 substrate and utilize PIN diodes to attain the reconfiguration. In this paper, a novel reconfigurable antenna operating in the 2, 4, and 6 GHz bands is presented. Reconfiguration is achieved using a single PIN diode. The antenna was simulated and implemented. The fabricated antenna exhibits three modes of operation with resonances at 2.07, 4.63, and 6.22 GHz. The three modes were defined by different values of the forward biasing voltage applied to the diode, in the range of just zero to 900 mV. The return loss is well below 40 dB for the primary resonance in every mode, and the VSWR is minimal, 1.01–1.02. The experimental results of the radiation patterns suggest that the gain is maximum at the bottom side of the antenna and equal to 5.48 dBi when operating in the 2 GHz band, 3.97 dBi on the bottom side when operating in the 4 GHz band, and 12.79 dBi in front of the antenna when operating in the 6 GHz band. If the antenna works for transmission or reception solely in the forward direction, it should be rotated toward the front by an elevation angle between 12° and 25° for maximum gain. In this case, depending on the rotation angle, the maximum gain which can be attained in the 2, 4, and 6 GHz band resonances is 3.81, 0.36, and 12.79 dBi, respectively. Finally, the fabricated antenna demonstrated better functionality in the 4 GHz band than suggested by the simulation, while the measured gain was much higher in all cases. Overall, the deviation between the simulation and the experimental results was negligible.


This paper presents a CoPlanar-Waveguide (CPW)-fed stingray-shaped Ultra-WideBand (UWB) Multiple-Input–Multiple-Output (MIMO) antenna system designed for microwave imaging applications. Featuring a diagonal square with four inner lines and a vertical line at the center from toe to tip with a CPW feed line, the unit antenna element looks like a stingray fish skeleton and is, therefore, named as a stingray-shaped antenna. It offers a bandwidth spanning from 3.8 to 12.7 GHz. Fabricated on a 31mil RO5880 RF teflon substrate with a relative permittivity of 2.2, the proposed antenna has dimensions of 26 × 29 × 0.787 mm Within the realm of wireless communication, ultra-wideband (UWB) antennas have garnered increasing recognition both for millimeter waves and sub-6-GHz systems ,2. UWB aSeveral UWB antenna systems are developed in the literature ,14,15,16In UWB MIMO systems, multiple antenna elements are utilized at both the transmitting and receiving sides to enhance communication reliability and capacity. These systems leverage spatial diversity and multipath propagation to effectively counter signal fading and interference, resulting in elevated data rates and enhanced resilience in wireless communication ,18. The hing. In , a two-phing. In , artifiching. In , a four-hing. In  presentehing. In , a UWB shing. In , a four-hing. In , a four-hing. In , a MIMO hing. In , a four-This research presents a novel CPW-fed stingray-shaped antenna for biomedical applications, especially for tumor detection in soft tissues. The antenna aims for wideband coverage with adequate time-domain characteristics crucial for precise microwave imaging. Experimental results demonstrate effective performance across a broad frequency range from 3.4 GHz to 18 GHz, with a maximum realized gain of 3.6 dBi at 7 GHz, enabling adequate microwave signal boost for capture and transmission. The antenna exhibits low group delay (<1.5 ns) for face-to-face and side-by-side configurations, minimizing pulse distortion critical for microwave imaging. A MIMO UWB antenna system with enhanced isolation (up to 17 dB) is developed. A comprehensive comparison with other UWB antennas in the literature highlights the CPW-fed stingray antenna’s unique advantages and improved performance, demonstrating its potential for practical microwave imaging applications.The performance of the antenna system has a critical role in the overall performance of the near-field microwave imaging systems. In addition to wide bandwidth, the antennas should be able to transmit pulses with little signal distortion. Also, the transient response of the antenna is critical to have proper penetration into the human tissue with minimal signal waveform distortion. This translates into a pulse duration in the order of nanoseconds, which means there is a multi-gigahertz bandwidth requirement for the antenna. The detailed requirements for antenna design specifications are given in . This woThe remaining part of this paper focuses on ultra-wideband antenna design for near-field microwave imaging and is organized into five sections. A = 10 mm, and B = 8 mm. A and B refer to the width of the vertical line and the diagonal line, respectively.The frequency domain performance of the antenna system has distinctive properties such as gain, polarization, and directivity. However, the specific performance requirements of the ultra-wideband antennas are derived from the time domain analysis, which gives important performance criteria for impulse-driven systems and determines if the system is compliant with the requirements of the intended application. The transient response gives insight into the angles of arrival and departure and polarization.R is the distance from the antenna, D is the maximum dimension of the antenna, and The time domain performance of an antenna system is analyzed in the far-field region where the antenna’s radiation pattern is independent of the distance from the antenna. In order to save the computation resource, the antennas should be placed where the far-field region starts. Equation  defines As the time domain responses show, there is a minimal distortion between face-to-face and side-by-side configurations. This corresponds to a high fidelity factor. The distortion between the radiated and captured signals is given with a cross-correlation value. The highest value of the cross-correlation is defined as the fidelity factor. The fidelity factor is contingent upon the system specifications. It is dependent on the performance parameters of the antenna and the waveform of the excitation signal. The antenna bandwidth should be wide enough to cover the entire pulse spectrum of the excitation signal so that the antenna distortion effect can be properly analyzed . The fidThe face-to-face configuration has a higher fidelity factor of −94.5 than the side-by-side configuration of 88.62. A better fidelity factor means lower distortion in the transmitted wave, which is an indicator of a wider bandwidth of the antenna system. The high-fidelity factor antennas are essential for microwave imaging applications. Another important performance parameter of the ultrawideband antennas is the group delay. Group delay is defined as the amount of change in the transmitted signal phase with respect to the frequency change as given in Equation (3)τ=−∂θIt varies across the operating spectrum of the antenna system. Microwave imaging systems require a group delay value of less than 2 ns . By asseeristics . Howevereristics ,36. The eristics . Figure The introduction of the isolating structure yields a remarkable enhancement in the isolation response. This enhancement is readily apparent when examining the port isolation values, with notable improvements observed. Depending on the antenna, the isolation values have improved from a minimum of 3 dB up to 20 dB. The isolating structure tremendously improved the isolation, especially between the adjacent antennas, which are antennas 1 and 2, antennas 2 and 3, antennas 3 and 4, and antennas 4 and 1.In It was an interesting observation when non-radiating elements showed noticeable electric currents on their surface in a simple MIMO structure. This was due to the coupling of electromagnetic fields from the radiating elements to the nearby non-radiating ones. This would manifest itself as a reduction in bandwidth and cause distortion. Nonetheless, after a series of experiments, a practical solution in the form of a decoupling structure is employed to reduce these interactions. Many types of simpler truncated polygon structures have been evaluated as decoupling elements. Finally, a unique isolating structure that was derived from the antenna itself is evaluated. The shape and even the length of its corner stubs had a significant effect on the isolation performance. None of the previous experiments were as good as this unique structure for isolating the MIMO antennas. Upon its integration, a noteworthy transformation became evident. Specifically, at 4.5 GHz b, the coSimilarly, at 7.4 GHz d, the cuThe proposed stingray MIMO antenna system is fabricated and tested using the in-house facility. The small differences between the isolation performance of the reciprocal ports of the MIMO antenna system can be attributed to calibration drift, measurement uncertainty, and measurement performance shifts caused by the small bends due to the antenna’s flexibility. Nonetheless, despite the difficulty of keeping these under control, the measured isolation performance is quite successful, reaching a minimum of 22 dB, as shown in xz-plane, by having some gain suppression at two locations. Meanwhile, in the yz-plane, the pattern has some level of gain suppression from all four directions. It is evident that radiation in the vertical direction is more efficient.The simulated and measured 2D radiation pattern of the proposed stingray MIMO antenna at two different frequencies of 6.7 GHz and 11 GHz are shown in xz-plane. This pattern ensures broad coverage in the horizontal plane, making it suitable for various communication and sensing applications. It is compulsory to highlight that the radiation pattern results present a good agreement between simulation and measurement data despite the mechanical challenges faced during the measurements. This agreement validates the accuracy and practicality of the design, proves the success of implementation, and confirms its performance over the distinct frequencies across the band.At higher frequencies, such as 11 GHz e, the anr is the cross-polarization ratio. It can be expressed as Equation (The MIMO performance parameters are shown in Equation .(5)r=10The This study presents a CPW-fed stingray-shaped UWB antenna element and its orthogonal-oriented four-port MIMO antenna implementation without a novel isolating structure. The single antenna element has a 26 mm × 29 mm while the MIMO structures have 58 mm × 58 mm size. By incorporating an isolating structure, the isolation performance is improved up to 16 dB. The simulation and experimental results exhibit a good agreement, affirming the accuracy of the simulations and the precision of the practical implementation. The measurement results show that the MIMO antenna had a bandwidth of 10 GHz covering a frequency range of between 4.4 GHz and 14.4 GHz. The overall gain of the antenna system is 5.3 dBi; the typical isolation between the antenna elements is 22 dB. Performance metrics pertinent to Multiple-Input–Multiple-Output (MIMO) systems, such as the Envelope Correlation Coefficient (ECC), Mean Effective Gain (MEG), Channel Capacity, and Diversity Gain (DG), are compliant with the requirements of the target applications. The ECC is less than 0.01, while the MEG is calculated between −3 and −4 for all antenna elements across the frequency. Furthermore, a thorough evaluation of the antenna systems in the time domain reveals that they are well below the acceptable thresholds. The proposed MIMO antenna design holds considerable promise for integration into MIMO-based medical imaging systems, where antenna efficiency plays a pivotal role. Elevated levels of isolation, with the proposed novel isolating structures, promise to enhance the precision of medical imaging significantly. The design also offers a powerful solution that can enhance the robustness of wireless connections. This enables smooth data transfer and meets the growing needs of modern consumer electronics. Exploiting the benefits inherent in UWB MIMO technology, this antenna system is positioned to substantially enhance the user experience, marking a new era in unlicensed wireless applications over a range of ultra-wideband frequencies. These applications cover various areas, including personal area network wireless connections, radar systems, microwave imaging, and beyond. The future scope of this work will be extended to a system with more MIMO elements integrated with a feeding network realized with cost-effective materials.


An ultra-wideband and high-gain Vivaldi antenna with artificial electromagnetic material, suitable for ground-penetrating radar (GPR) systems, is proposed. Directors loaded inside the antenna gradient slot direct electromagnetic waves by inducing current to improve gain. The artificial electromagnetic material, also called metamaterial, is composed of multiple “H”-shaped units arranged in a certain regular pattern, loaded at the antenna aperture. The artificial electromagnetic units affect the antenna radiation waves by changing the refractive index to improve radiation directivity. The four Vivaldi units are arranged into a horn-shaped array, and each two units are orthogonally fed to realize dual polarization. Experimental results demonstrate that the antenna has good impedance matching of  The demand for GPR technology is becoming increasingly urgent due to the growing requirement for underground resource development and underground target detection. GPR is widely used in the detection of hidden targets such as underground cavities, bridge defects, underground pipelines, and archaeology, due to its advantages of being non-destructive, fast and high resolution. GPR uses the penetrating property of electromagnetic waves to detect targets underground and behind walls. The higher the frequency of the electromagnetic wave, the higher the loss when penetrating the ground and walls. Thus, a GPR system mostly uses a lower operating frequency . In ordeGround-penetrating radar antennas not only need to meet the requirements of UWB, but also require strong directivity. The directivity of an antenna refers to its ability to radiate and receive differently in different directions. Ground-penetrating radar needs to detect underground targets, and in order to minimize interference, high-directional antennas should be used . AntennaUp to now, there have been many methods that improve the bandwidth and gain of the UWB antenna for a GPR system. For example, substrate-integrated waveguide (SIW) is loaded onto a dielectric substrate . SIW is In recent years, artificial electromagnetic material, known as metamaterial, has been widely used to enhance antenna directivity, improve antenna gain and concentrate the antenna beam. Artificial electromagnetic material is formed by the periodic or aperiodic arrangement of artificial units whose sizes are sub-wavelength, which can be equivalent to real electromagnetic material in a macro sense. Theoretically, it can achieve an arbitrary dielectric constant and permeability, such as zero refractive index and negative refractive index . MetamatA four-element director is loaded into the gradient slot of the Vivaldi antenna unit, and the electromagnetic wave is directed to the edge of the director to enhance the middle and high frequency gain;“H”-shaped metamaterial units are designed and loaded into the aperture of the Vivaldi antenna in a periodic arrangement to make the beam more concentrated by changing the refractive index.In this paper, based on the shared aperture dual-polarized Vivaldi antenna, in order to further improve the gain, the following two methods are adopted:By combining all of the above methods, the gain can be increased to 15.2 dBi without affecting the bandwidth, and the working frequency is between 0.9 and 4 GHz.The configuration of the high-gain dual-polarization Vivaldi antenna is shown in A quadruple director with a size of approximately The antenna designed in this paper can maintain a good return loss at 0.9–4 GHz, lower than −10 dB, and achieve a gain of 15.2 dBi by loading the directors and metamaterials.The director is loaded inside the tapered slot of the initial Vivaldi antenna, which increases the gain of the antenna without changing its size. When the Vivaldi antenna is fed, the radiation electric field produced by the antenna points from one side of the tapered slot to the other side, that is, the radiation electric field is along the x-axis. The electric field of the antenna obtained by CST Studio Suite (CST) software simulation is shown in When the Vivaldi antenna operates, the feeding current is located between the tapered slot lines and propagates toward the aperture along the slot lines. Due to the short distance between the slot lines near the feeding end, which is less than half a wavelength of the antenna’s operating frequency, the electric field transmits toward the antenna aperture within the substrate without radiating outward. As the current travels, when it reaches the slot line with a spacing of half the operating wavelength of the antenna, electromagnetic waves begin to radiate. At this point, the antenna can be equivalent to a radiation source, generating radiation electromagnetic waves. When the length of the director is shorter than half the operating wavelength, the induced electromagnetic waves generated by the induced current of the director are phase-delayed compared to the radiation electromagnetic waves, thereby directing the radiation. Conversely, when the length of the director is longer than half the operating wavelength, the induced electromagnetic waves generated by the director are phase-advanced and will reflect the radiation. The position of the director also affects the antenna’s performance. When the length of the director is less than half the operating wavelength and is located close to the radiation source, it plays a role in directing the radiation. However, when the length of the director is too far away, it will reflect the radiation electromagnetic waves, thereby degrading the antenna’s gain.Based on the above principles, the influence of different lengths of the directors on the antenna gain are compared, as shown in When electromagnetic waves propagate in the air and enter the medium at a certain angle, according to Snell’s law of refraction:n1sinθ1=nThis article presents the design of periodic H-shaped metal patches printed on a 1.5 mm thick Rogers RT5880 substrate. n is the effective refractive index, k is the free space wave number, d is the effective substrate thickness, z is the wave impedance. Since the S-parameters are complex, the obtained effective refractive index and wave impedance are also complex. The imaginary part of the refractive index represents the absorption of the electromagnetic wave by the medium, while the real part of the wave impedance represents the attenuation of the electromagnetic wave. Since the medium is a passive medium, the imaginary part of the effective refractive index must satisfy According to the formula proposed in reference :3)n=1kd12+S212,z=1+S112−n=1kd12+SThe square root of a complex number has two values: m is the number of periods. By substituting the simulated S-parameters into Formula (Formulas  and 6)  6) can bn=−jkdlnΔIt can be seen that the “H”-shaped artificial electromagnetic material undergoes resonance at around 13 GHz and maintains a high stability in the frequency range of 0.5–6 GHz. For the required frequency band of 0.9–4 GHz, the “H”-shaped artificial electromagnetic material is in a non-resonant state, and the imaginary part of the effective refractive index is about 0, while the real part gradually increases from 1.68 to 1.78. Loading the metamaterials in front of the Vivaldi antenna aperture can be used to focus the beam and to improve the gain. The gain improvement effect obtained through simulation is shown in The simulation results show that the maximum gain of the Vivaldi antenna unit with metamaterials is increased by approximately 1 dB. The results indicate that the artificial electromagnetic material can significantly improve the high-frequency gain, but the effect is not ideal at low-frequency. This is because the artificial electromagnetic material is a sub-wavelength artificial structure, and in order to achieve the same gain improvement effect at lower frequencies, a larger structure is required. Considering the size of the antenna, it was decided to use the dimensions of the lens structure mentioned above.As the designed Vivaldi antenna requires four-port feeding, a wideband Wilkinson power divider  was desiThe substrate used for the designed wideband one-to-two power divider in this article is Rogers RT5880, with a thickness of 0.508 mm, relative permittivity of 2.2, and a size of 54.3 × 40 mmThe proposed antenna array was fabricated, and the performance was measured using the vector network analyzer (E5063A) and the microwave anechoic chamber. The four Vivaldi units were fixed together using screws and were connected using conductive metal stickers, and the antenna prototype is shown in The reflection coefficients obtained through simulation and vector network analyzer measurements are shown in The radiation patterns and gain of the antenna were measured using the microwave anechoic chamber and were compared to the simulated results. This paper proposes an ultra-wideband and high-gain Vivaldi antenna for a ground-penetrating radar system. The gain is enhanced by loading directors inside the gradient slot of the Vivaldi antenna and by periodically arranging artificial electromagnetic material in front of the antenna aperture. The four ports of the antenna are connected to Wilkinson power dividers to obtain high port isolation and are fed orthogonally to form dual polarization. The antenna has good impedance matching throughout the entire operating frequency band of 0.9 to 4 GHz, and the realized gain can reach between 6.7 to 15.2 dBi. The resulting antenna is electrically large, with dimensions of 0.72 × 0.72 × 0.86


In this paper, a 38 GHz 4-port multiple-input multiple-output (MIMO) antenna with considerable isolation and gain enhancement for 5G applications is introduced. The suggested antenna element is a monopole antenna composed of a circular patch with a rectangular slot etched from it and a partial ground plane is used to extend the desired frequency to operate from 36.6 GHz to 39.5 GHz with a center frequency of 38 GHz. The high isolation is achieved by arranging the four elements orthogonally and adding four stubs to reduce mutual coupling between elements at the desired frequency bands. The gain improvement is also introduced by placing a frequency selective structure (FSS) which is designed at the same frequency bands of the antenna under the suggested MIMO antenna to act as a reflector. The proposed four-element MIMO with the FSS prototype is built and tested in order to confirm the simulated results. The suggested antenna operated from 37.2 GHz to 39.2 GHz with an isolation of less than 25 dB across the obtained frequency range. The peak gain of the antenna is enhanced from 5.5 dBi to around 10 dBi by utilizing the FSS structure; furthermore, the back radiation is enhanced. The MIMO performance is validated by extracting its parameters and comparing with the simulated results. The results extracted from the simulation and the measurement show satisfactory matching along with the target band, indicating that the proposed structure could be used for 5G communications. The fifth-generation 5G) communications are distinguished by three distinct characteristics: universal connectivity, extremely low latency, and extraordinarily high data transmission rates ,2. This G communiAntenna design is one of the most complicated issues for future 5G cellular connectivity. Various experts have been working on 5G antennas that resonate at a frequency of 38 GHz ,11,12,13Gain enhancement can be accomplished by utilizing several techniques such as employing array configuration ,19,20, aA feasible configuration of a slotted bowtie antenna with an AMC structure is introduced in . In 22]22], a hi3 are joined and positioned orthogonally with four stubs. The MIMO testing findings in terms of impedance and radiation characteristics are extracted to investigate the desired performance of the MIMO antenna. Moreover, the MIMO diversity parameters such as envelope correlation coefficient (ECC), diversity gain (DG), channel capacity loss (CCL) are also extracted. The novelty of this work is the design of a simple 4-port antenna operated at 38 GHz applications. Second, the antenna achieved isolation between ports around more than 25 dB, which is suitable for this application. Third, the antenna has an enhanced gain of 10 dBi with the help of the FSS structures. Fourth, the antenna has a suitable overall size and diversity performance which is recommended for 38 GHz applications.In this paper, a 38 GHz MIMO antenna composed of highly isolated four elements with an FSS structure is designed and simulated using HFSS for 5G applications. To achieve the anticipated 5G frequency ranges, the circular patch is cut by a rectangular slot. Furthermore, to accomplish the high isolation properties of the MIMO configuration, the four elements of the recommended antenna with a size of 25.95 × 25.95 × 0.238 mmr = 3.55, with an overall size of L × L = 12 × 12 mm2. First, a circular patch monopole antenna is designed with a diameter R = 4.94 mm, a partial ground plane with a length (Lg1) of 8 mm, and a 50 Ω feedline with a width (Wf) of 0.4 mm and a length (Lf) of 7 mm is introduced as a start point of the design. As shown in The single antenna design phases are shown in 11 level while the bandwidth of the antenna was not affected. The 2D layout with the optimized dimensions is shown in 11 outcomes are displayed in 11 levels of −30 around 38 GHz in the recommended band.From the design procedures above, antenna 3 is recommended for producing the desired 38 GHz frequency bands. By using the HFSS simulator, a parametric study is performed to obtain the optimized width (W1) of the rectangular slot as shown in 2. The single antenna unit discussed in the previous section is copied three times and placed orthogonally to each other as depicted in This section discusses the structure of a 4-port MIMO antenna and the method used to enhance isolation between elements. As indicated in 11 with and without stubs shows that the two antennas operated at approximately the same frequency bands. The return loss between antenna elements (S21/S31/S41) is improved by around 5 dB by introducing the four stubs, especially at 38 GHz, which validates the MIMO antenna with stubs to be used instead of without stubs in this paper.For achieving the high performance of the four-element MIMO antenna, parametric studies were performed. The parametric study to show the influence of changing the length of isolation stubs (Ls) on antenna performance operates as depicted in 3 and is joined and positioned orthogonally with four stubs.11 reaches −24 dB, and the coupling between antenna elements (S21/S31/S41) is <−22 dB. On the other hand, the tested results which are extracted using (R&S ZVA 67 VNA) are from 37 GHz to 39 GHz with S11 reaching −36.2 dB, and the isolation between ports (S21/S31/S41) is <25 dB. There is a slight difference between the two outcomes due to fabrication and measurement tolerances that cannot be resolved.This section investigates the characteristic of the FSS unit cell and studies the effect of the cell size on the MIMO antenna performance.a. The FSS Unit cellr = 2.2, and a total size of 2.82 mm × 2.82 mm are utilized in the simulation and fabrication. A copper layer of 0.035 mm in thickness with two rectangular slots is added on top of the substrate, as shown in 21 response lower than −10 dB, and the lowest level is introduced at 38 GHz. Additionally, S11 touches 0 dB which means the FSS can be used as a reflector.As discussed in the literature review, The FSS is considered one of the techniques utilized to improve the antenna gain. By placing the FSS array under the antenna, it can be worked as a reflection structure to reflect the back-radiation and enhance the radiation characteristics of the antenna. The FSS unit cell and the suggested FSS array are shown in b. Antenna attached with FSS array11 and peak gain, the suggested MIMO antenna is attached with different FSS cells size and placed at a distance of 5 mm, as shown in To investigate the effect of the FSS cells array on the MIMO antenna performance, such as Sr = 1.03 of polystyrene and 5 mm thickness is added between the MIMO antenna and the FSS cells, as shown in 11 and isolation between elements (S21/S31/S41) are shown in Depending on the previous discussion to improve the suggested antenna gain, a 14 × 14 FSS array (36.2 mm × 36.2 mm) is added under the proposed MIMO. The suggested 4-port MIMO antenna with a 14 × 14 FSS cell is fabricated as shown in 11 reaching −21 dB, and the isolation between antenna elements (S21/S31/S41) is <20 dB. While the tested outcomes are accomplished by frequency bands from 37.2 GHz to 39.2 GHz, S11 reaches the maximum level of −24.4 dB at 38 GHz, and the isolation between antenna elements (S21/S31/S41) equals 25 dB. The simulated and measured outcomes have a good match within the operating band (38 GHz). However, due to the fabrication and measurement tolerances, there is a difference between the two results outcomes.The simulation result shows that the proposed antenna cover the band from 36.7 GHz to 39.5 GHz, with SThe normalized simulated radiation patterns outcomes of the 4-port MIMO antenna at port 1 at 38 GHz with/without FSS are illustrated in The normalized radiation pattern outcomes of the 4-port MIMO antenna with FSS at port 1 at 38 GHz are shown in The measured gain and radiation patterns are extracted by the technique discussed in ,31. It iThe diversity parameters of the MIMO antenna with FSS such as ECC, DG, and CCL are measured to determine the performance of the antenna in the MIMO system. The ECC can be accounted for using S-parameters and radiation field patterns to quantify the multiple port efficiency .(1)ECC=An ECC < 0.5 is considered to be a good level for MIMO systems . Figure The DG can be calculated from ECC using Equation (3) .(3)DG=1The CCL indicates the upper constraint on data transmission rate and is studied to show its effect on the MIMO performance. The CCL should be <0.4 bits/s/Hz . The CCLFor 5G communications, a four-element MIMO antenna with an FSS has been proposed. The proposed MIMO antenna was intended to be operated at frequency ranges from 37.2 GHz to 39.2 GHz, with isolation greater than 25 dB at the operating band. Gain enhancement has been achieved by employing an FSS compared to the MIMO configuration without an FSS structure. MIMO metrics such as ECC, DG, and CCL have been calculated from simulated and measured data to validate the diversity performance of the proposed antenna and to demonstrate its superior features. The simulated and measured data match well through the operated band, implying that the proposed structure can be recommended to be utilized in 5G communications.