A low-profile, wideband, and high-gain antenna array, based on a novel double-H-shaped slot microstrip patch radiating element and robust against high temperature variations, is proposed in this work. The antenna element was designed to operate in the frequency range between 12 GHz and 18.25 GHz, with a 41.3% fractional bandwidth (FBW) and an obtained peak gain equal to 10.2 dBi. The planar array, characterized by a feed network with a flexible 1 to 16 power divider, comprised 4 × 4 antenna elements and generated a pattern with a peak gain of 19.1 dBi at 15.5 GHz. An antenna array prototype was fabricated, and the measurements showed good agreement with the numerical simulations as the manufactured antenna operated in the range of 11.4–17 GHz, with a 39.4% FBW, and the peak gain at 15.5 GHz was 18.7 dBi. The high-temperature simulated and experimental results, performed in a temperature chamber, demonstrated that the array performance was stable in a wide temperature range, from −50 °C to 150 °C. An antenna is, by definition, a sensor of EM (electromagnetic) waves; it plays an extremely important role in radar sensor , biomediMany effective methods have been proposed in the literature to increase the operation bandwidth of microstrip patch antennas. Among them, the use of parasitic patches has been considered as well as optimization of the antenna shape. More specifically, a patch antenna adopting a U-shaped slot-fed and stacked structure was proposed in , reachinOther methods have been investigated to improve the gain of the antenna element. In ,22,23, tThis study proposes a novel double-H-shaped slot microstrip patch antenna, robust against high temperature variations, as a radiating element of a low-profile, wideband, and high-gain patch antenna array. In order to test the effectiveness of the proposed design, the antenna was fabricated and the prototype measured for comparison with the numerical results. The analysis of the temperature resistance was performed in a temperature chamber between −50 °C and 150 °C.Accordingly, the remainder of this article is as follows. The elements of the antenna and antenna array designs are presented in rε = 3.66, tan δ = 0.004), and a ground plane  with a double-H slot fed through a microstrip line. The second part is a coplanar radiator, which includes four parasitic patches, a ground plane  connected with the ground plane of the first part by means of metallic pin columns, and a dielectric substrate . The third part, filling the space between the two radiators, consists of air (or foam).The proposed antenna element is shown in The antenna element proposed in this paper uses a novel double-H-shaped fed slot, which allows for the generation of multiple resonant modes thanks to the adopted double patch structure. More specifically, the coplanar parasitic radiator and the patch radiation structure form a resonant cavity, which reduces the Q-factor and increases the antenna’s bandwidth. As shown in In the first stage (Stage I), a patch antenna with a single H-shaped slot was considered. The resonant frequency of the microstrip patch working in the In the second stage (Stage II), two H-shaped feeding slots were considered in the ground plane. The geometrical parameters of the slot  satisfie17.7 GHz a, with 717.7 GHz b.The third stage (Stage III) showed that the antenna’s bandwidth was greatly improved by adopting a stacked structure. Moreover, the peak gain increased to 8.8 dBi. The distance The last stage (Stage IV) consisted of the final design, with a bandwidth of 12 GHz to 18.25 GHz (41.3% FBW) and a peak gain of 10.2 dBi. x-axis and worked at different frequency bands. For Mode 1, the bandwidth of |MS| > 0.707 was between 12 GHz to 14.7 GHz, which can be considered as the lowest frequency part of the bandwidth in combination with the operating frequency band of the previous feed antenna. For Mode 2, the bandwidth of |MS| > 0.707 was 14.7 GHz to 20 GHz. By stimulating these two modes, the antenna can achieve a high gain in a wide band. In mode 3, the bandwidth of |MS| > 0.707 was between 14.5–20 GHz. The current of the parasitic patches and ground 2 was in the opposite direction along the x-axis and in the offset state. When using this mode, one needs to enhance or weaken a certain mode according to the mode of the patch radiator. The current components of the parasitic patch in mode 4 along the x and y axes were in the offset state, and it could not be used as the main radiation mode. Modes 5, 6, and 7 were unsuitable for linear polarization and high gain. The combination of modes 1, 2, and 3 of the coplanar radiator and the patch radiator enabled the overall structure to obtain a high gain and a high bandwidth.To maintain high gain over a wide bandwidth, the characteristic mode of the coplanar radiator was analyzed using HFSS to exploit the mode in order to generate a pattern with high gain. Lp  was properly tuned. As is evident from Lp = 12.5 mm.Indeed, the antenna resonated in the frequency range between 12 GHz and 18.25 GHz, with a 41.3% FBW, and the peak gain was equal to 10.2 dBi at 15.5 GHz. In order to achieve such a performance, the value of the parameter The working principle of the proposed antenna element can be described as follows. At the frequency f = 13 GHz, the radiation patch worked as shown in For the sake of comparison, In addition to the antenna element, the design of a 4 × 4 antenna array, shown in S11| values below −15 dB from 9 to 20 GHz. As for the array, the values of the simulated reflection coefficient and of the peak gain given in S11| < −10 dB) from 11.4 GHz to 17 GHz, and the maximum gain was 19.1 dBi.The simulated reflection and transmission coefficients at positions 1, 2, and 3 of The prototype of the proposed 4 In order to verify the performance of the proposed antenna in high-temperature environments, the measurement of the reflection coefficient was carried out in a high–low temperature chamber, as shown in More specifically, an ESPEC (SH-642) high–low temperature chamber and a vector network analyzer were used. The measurement considered nine different temperatures, namely, 20 °C, −20 °C, 0 °C, 40 °C, 60 °C, 80 °C, 100 °C, 130 °C, and 150 °C. The behavior of the magnitude of the reflection coefficient versus the frequency is shown in The values of the minimum frequency (M1), the maximum frequency (M2), and the bandwidth of the curves shown in According to the datasheet on the substrate material, the expected variation of the permittivity in the temperature range between −50 °C and 150 °C was about 3.66 ± 0.05. Therefore, the impact on the magnitude of the reflection coefficients was also simulated . Like inMoreover, although the power pattern could not be measured in the high–low temperature chamber, it has been simulated when changing the material property, and the results in The design of a low-profile, wide-band, and high-gain 4 × 4 antenna array based on a novel double-H-shaped slot microstrip patch radiating element, robust against high temperature variations, was presented. The corresponding prototype was fabricated, and the measurement results demonstrated that the antenna array reached a 5.6 GHz bandwidth  and a maximum gain of 18.7 dBi. The test and simulation results also indicate that the proposed antenna array had good stability in a temperature range from −50 °C to 150 °C, and that, therefore, the antenna has good potential to be applied in higher-temperature environments.


This paper describes the design steps carried out to prove the concept of a wideband monopole antenna system to be used in a wearable device conceived for the evaluation of electromagnetic field radiation. Such a device is envisaged to be integrated into protective vests worn by professional users in their working space environment as part of intelligent multi-risk protection. Initially, the main characteristics of a simple straight monopole are reviewed to serve as a reference. A modified octagonal monopole antenna element is introduced, and a two dual-linearly polarized configuration of such monopoles is designed, fabricated, and tested to be used in the frequency range of 0.7–3.5 GHz. The expected radiation characteristics  are confirmed experimentally. The effects of a thick lossy foam substrate layer used to mitigate the presence of the metal shield, employed in the vest lining as a Faraday cage protection, are analyzed both by simulation and experimentally. Finally, electromagnetic simulations are carried out to confirm that a system of five dual-linearly polarized monopole elements located in the chest, shoulders, back, and helmet of the user can provide an adequate estimation of the incident electromagnetic field radiation. In the last thirty years, there has been a remarkable increase in the deployment of mobile communication and Wi-Fi systems. Notably, 2G, 3G, and 4G standards represent a step forward in such an increase. With the advent of 5G, an enormous boost in the number of users and terminals, as well as in traffic, is expected . With thThis paper presents the proof-of-concept for the development of an antenna system envisaged to be integrated into protective vests worn by professional users in their working space environment as part of intelligent multi-risk protection . Such a The proof-of-concept for the development of the antenna system is divided into three steps. The first step corresponds to the design and optimization of the basic antenna element, a wideband dual-linearly polarized monopole, in free space . In the The modification was introduced in the octagonal printed planar monopole antenna element to further meander the current distribution.The compact combination of two linearly polarized monopoles was used to obtain dual-linear polarization, with acceptable mutual coupling.A foam absorber layer was used to mitigate the unwanted effects of a metal shield.The configuration of the antenna system was used to provide full coverage of the environment around the user.A method was used to estimate the amplitude of the electric field of the incident electromagnetic plane wave.It is important to highlight the following innovative features proposed and described in this paper:The paper is organized into five sections. After this introduction , SectionThe dipole antenna has been used since the 1887 Heinrich Hertz founding experiment to prove the existence of electromagnetic waves predicted by James Clerk Maxwell . It is oThe simplest printed monopole antenna configuration is shown in It is composed of a straight metal strip printed on one side of the substrate and the ground plane printed on the other side. This simple monopole will not fulfill the impedance bandwidth specifications (0.7–3.5 GHz). It is presented and analyzed here just to serve as a reference.s = Ws = Wg = 150 mm, Lg = 50 mm, and Wm = Wf = 3 mm. A cheap 1.6 mm thick FR4 substrate with relative permittivity εr = 4.3 and loss tangent tanδ = 0.025 was chosen. The monopole length was optimized to have the first resonance at the central frequency (f0 = 2.1 GHz) of the working frequency band 0.7–3.5 GHz. Lm = 25 mm = 0.175λ0 was obtained. The microstrip feed line width was swept from 3.0 mm to 0.5 mm with a 0.5 mm step. This corresponds to characteristic impedances of the microstrip feed line ranging from 50.7 Ω up to 110.6 Ω [Initially, the following dimensions have been used: L 110.6 Ω . The inp 110.6 Ω  are show11| = −10 dB reference level, which corresponds to a 10% reflected power level. With this criterion, the larger bandwidth is obtained for Wf = 1.0 mm and is about 46% (1.78–2.86 GHz), far from the required 133.3% or 5:1 (0.7–3.5 GHz), as expected.We define the impedance bandwidth using the usual |Sf = 1.0 mm configuration. The results obtained for the 2D cuts in the principal planes are shown in θ polarization component is meaningful. Especially in the H-plane, for 1.7 GHz and 2.9 GHz, the amplitude of the Eφ component is similar to the Eθ one, for φ ≈ 30° . This is a specificity of the planar monopole that can be explained by the current distribution in the ground plane  is specified, a simple configuration that meets the requirements was selected. Initially, the octagonal-shaped monopole ,31 was si ≠ Lvh) to increase flexibility in the design process. The values used for the geometrical parameters are indicated in A simple rectangular ground plane and a thin single-layer substrate configuration are used. Moreover, a coplanar waveguide (CPW) feed is utilized. The novelty of the proposed configuration corresponds to the use of right triangles on four edges of the octagonal patch to enhance meandering. The octagonal patch is not regular  because the other geometrical parameters have been optimized for that case. It can also be concluded that the design equation of the octagonal monopole applies, that is, the patch length is slightly less than a quarter of the free-space wavelength at the frequency of the first resonance  [It can be concluded that for the two ground plane widths considered, the ground plane width has a small effect on |S 100 mm) ,31.The simulated current distribution on the modified octagonal monopole, at the limits and center of the frequency band of operation, is shown in The simulated far-field 2D radiation pattern results are shown in The YZ plane radiation pattern does not change much in the whole frequency range of operation. However, on the other two principal planes, the radiation pattern is quite different at 3.5 GHz. This difference can be justified by the existence of the higher-order modes already referred to above ,41. The The gain simulated results are shown in To avoid the possibility of a complete polarization mismatch, two modified octagonal monopoles with orthogonal orientations were combined, as shown in The amplitude of the experimental S-parameters is shown in This section deals with the dual-linearly polarized monopole antenna element in the presence of a metal shield . Lossy foam is placed between the substrate, the ground plane, and the metal shield to absorb the reflections. First, the foam material is macroscopically characterized experimentally, and then the foam layer thickness is optimized.0), and therefore only ε and σ need to be measured.The absorber layer was obtained by cutting slices of the planar region of an anechoic chamber absorber panel. As the macroscopic characteristics  of the panel material are not homogenous, they need to be measured. However, the foam has normal magnetic behavior  relative electric permittivity are defined asn method , where tr′ and εr″ experimental results, obtained for part of the frequency band of interest, are shown in As an example, εThe dual-linearly polarized monopole antenna element with an absorbing foam layer and a metal shield is shown in This antenna structure has been simulated using an absorbing layer with thicknesses of 5, 10, and 15 mm. The obtained amplitudes of the S-parameters are shown in 11| (and |S22|) below −10 dB except in the range of 0.70–0.75 GHz where it is slightly above (−7.3 dB at 0.7 GHz). In addition, for an absorber thickness of 10 mm, the mutual coupling is below −14.2 dB.An absorber thickness of 10 mm provides |S11| and |S21| experimental results are shown in 22 results (not included) generally show a good agreement with S11. The experimental results of the free-space prototype are also shown for reference. It can be verified that |S11| < −15.4 dB, and |S21| < −17.1 dB for the whole frequency band of operation. These results are substantially below (about 5 dB on average) the simulated one, especially at the lower frequencies. This is due to the less accurate characterization of the absorber in that frequency region as a pair of horn antennas is not available for such a frequency band. It can be concluded that it is feasible to use an absorber layer a few millimeters thinner.An antenna prototype with a 10 mm thick absorber layer has been fabricated and tested. The corresponding |SThe simulated radiation pattern results for each port are shown in As intended, the metal shield almost blocks the radiation to its back side. There are several physical symmetries reflected in the radiation pattern. For instance, the theta component of Port 1 in the XY plane corresponds to the phi component of Port 2 in the XZ plane, rotated 90 degrees, and vice versa. In the three principal planes, the radiation intensity level in the front hemisphere, for both polarizations, is below—5 dB in only one-third of the 180-degree angular region. Then, it can be concluded that the radiation pattern does not change much in most of the front hemispheres.The corresponding realized gain and radiation efficiency simulated results are shown in The gain spans the range of −3.4 to 0.3 dBi with an average of −1.7 dBi. Compared with the gain of the free-space monopole shown in  there isThe radiation efficiency decreases as frequency increases and spans the range of −9.9 dB (10.7%) at 3.5 GHz to −6.3 dB (23.6%) at 0.7 GHz. The decrease in efficiency is a natural consequence of the insertion of the absorber layer, a price to pay for the mitigation of the presence of the metal shield.The dual-linearly polarized monopole antenna element, analyzed in the previous section, is used as the basic antenna element of the envisaged antenna system used to evaluate the incident electromagnetic field.The antenna system proposed for evaluating the incident electromagnetic field is shown in Five dual-linearly polarized monopole antenna elements are spatially distributed to provide almost uniform coverage of the space around the user. Four such antennas are located on the torso , and the remaining one is located on the helmet. The torso has maximal dimensions 40 cm × 40 cm× 25 cm This simple model of the user was chosen according to the facilities that will be used to fabricate the physical phantom  needed tThe proposed antenna system was simulated using CST Studio Suite. The input reflection coefficient and gain of each antenna port have been obtained. Moreover, in receive mode, the voltage induced in each antenna port is also obtained for different directions of incidence and polarizations of an incoming plane electromagnetic wave. The port numbering is indicated in 77| = −7.1 dB at 3.5 GHz. As for the results shown in The simulated input reflection coefficient results are shown in To simulate the voltage induced in each port by the electromagnetic radiation present in the environment, a plane electromagnetic wave, with an electric field intensity of 1 V/m, was considered. The direction of incidence was varied in the angles θ (from 0 to 180 degrees) and φ (from 0 to 360 degrees) in 45 degrees steps. Vertical, horizontal, and 45 degrees inclined linear polarizations were studied. In the simulations, each of the 10 monopole ports was terminated with 50 Ω loads, which corresponds to the ideal situation of a perfect impedance match between the receiving antenna and the receiver.As an example, the incidence from θ = 90 degrees to φ = 0 of the plane electromagnetic wave with horizontal polarization is shown in As expected, Port 5 and Port 10, corresponding to the monopoles with line of sight (LoS) and the perfect (Z) polarization match, have the highest induced voltages. The induced voltage curves tend to follow the gain curve shown in 0) can be obtained from [e is the antenna effective height, and Cp is the antenna polarization mismatch factor. Assuming a perfect antenna impedance matching the voltage over the 50 Ω load (Vload) is half of V0. It is also known that [a = 50 Ω is the antenna input resistance, and Z0 = 120π Ω is the characteristic impedance of the vacuum. Combining Equations (2) and (3), and assuming Cp = 1, the incident electric field amplitude can be estimated as0 is the light speed in vacuum.The equivalent Thevenin-induced unloaded antenna voltage (Vned from (2)V0=E own that (3)he=λ Read the voltage at each of the 10 monopole antenna ports and select the highest one. Users will be instructed to move around while reading the voltages to avoid the eventual incidence of local radiation pattern minima.Obtain the monopole gain from est.Use Equation (4) to calculate EThe simplest procedure that can be used to estimate the amplitude of the electric field of the incident plane electromagnetic wave can be summarized as follows:It is assumed that there is a filtering device that selects the frequency band for each measurement. Taking into account the simplifying assumptions used, this simple procedure underestimates the amplitude of the incident electric field. A more accurate estimation can be obtained by implementing a more elaborate procedure and making use of the information available. For instance, we may use the amplitude and phase of the voltages measured at each port, obtain the direction of arrival (DoA) of the incident electromagnetic wave , and useAn antenna system based on a wideband modified planar printed octagonal monopole is proposed for integration into a system conceived for the estimation of electromagnetic field radiation in the frequency range of 0.7–3.5 GHz. This frequency range encompasses all the relevant mobile communication standards, namely, 2G, 3G, 4G, and most of the 5G (below 6 GHz) frequency bands. Such a system is envisaged to be integrated into protective vests worn by professional users in their working space environment as part of an intelligent multi-risk protection system.Initially, the simple straight monopole is completely characterized to serve as a reference. Then, the modified octagonal monopole with CPW feed is proposed to provide the required wideband. A dual-linearly polarized combination of such modified octagonal monopoles is proposed. The experimental macroscopic characterization of a lossy foam is described. A relatively thick layer of this foam is used to mitigate the unwanted effects of the metal shield on the dual-linearly polarized monopole antenna.Five dual-linearly polarized monopole antennas will be integrated into the user jacket, located on the chest, back, left, and right shoulders and helmet. Based on the knowledge of the monopole antenna gain, the frequency band of the operation, and the maximal induced voltage at the antenna ports, a simple procedure is proposed to estimate the amplitude of the electric field of the incident electromagnetic plane wave. As part of the tradeoff between the complexity and accuracy of the estimation procedure, other strategies may be followed.Use adequate wearable-oriented materials  .Integration into the vest .Tests with a realistic phantom .Uncertainty analysis of the procedure to estimate the amplitude of the electric field of the incident electromagnetic field.Extend the analysis up to 6.0 GHz (to include the remaining sub−6 GHz 5G frequency bands).Only the proof-of-concept of the antenna system is described in this paper. Therefore, no strict fulfillment of the specifications was imposed, as this is a task to be accomplished in the final stage of the development of the prototype by an industrial partner. Many topics of future work can be represented as requiring specific analysis for the evolution from the proof-of-concept to the industrial prototype, such as the following:We are completely aware that alternatives to the single modified octagonal monopole antenna element could have been adopted initially . However, we are convinced that using these other monopole geometries in the final antenna system would lead to an equivalent overall performance.


The outcomes of estimated and observed values are examined for MIMO inclusion factors such as DG, ECC, CCL, and MEG. The antenna’s performances, including radiation efficiency and gain, are remarkable for this antenna design. The designed antenna is successfully tested in a cutting-edge laboratory. The measured outcomes are quite similar to the modeled outcomes. This antenna is ideal for WLAN and Wi-Max applications.In this article, a 4 × 4 miniaturized UWB-MIMO antenna with reduced isolation is designed and analyzed using a unique methodology known as characteristic mode analysis. To minimize the antenna’s physical size and to improve the isolation, an arrangement of four symmetrical radiating elements is positioned orthogonally. The antenna dimension is 40 mm × 40 mm (0.42λ It is desirable to employ this modern technology, known as Ultra-wideband communication devices, to address the needs of high data rates at low costs. Since the Federal Communications Commission (FCC) approved the unlicensed 3.1 GHz–10.6 GHz range for UWB applications; it has evolved into a well-known innovation in the wireless communication industry. Because they constitute an essential part of UWB communication systems, UWB antennas have consequently generated enormous scholarly and scientific attention in recent years . To overIntroducing more components causes mutual coupling to rise, which lowers MIMO’s performance. Since electromagnetic interaction increases when MIMO units are put closely together and lowers MIMO performance, low mutual coupling is a crucial MIMO component. These techniques are used to initiate reducing mutual coupling: (1) an improper ground structure, (2) networks for decoupling, (3) parasitic components, 4) electromagnetic band gap (EBG), 5) lines of neutralization, and 6) meta material. To fulfill the expectations of the Internet of Things for more bandwidth and data rates, researchers have started to create UWB antennas to attain a high level of isolation in MIMO Antennas [ electrom lines of meta materations . A 4 × 4erations .3, and each radiator has a U-shaped slot. Excellent isolation and diversity measurements were attained [The 4 × 4 small antennas have 25 mm × 50 mm of surface area, and PIN diodes were utilized to turn on each radiator in the set up. An LC-shaped decoupling stub was employed for high isolation 2–12 GHz) across the band  GHz acro. A tiny attained  as a resattained . A two-pattained . A new tattained . A compaattained . To achiattained ,23,24,25But the above methods are not able to enhance the isolation. Therefore, we need an advanced antenna design approach that offers us great isolation and good diversity performance. A method of evaluation that is step-by-step is characteristic mode analysis. This design method is used to analyze antenna properties without using feed. The antenna structure is physically examined using CMA.The following sections compose the remaining document: In 0 × 0.44λ0 × 0.0176λ0), placed on a Fr-4 material with loss tangent of 0.002 and permittivity of 4.3. The Computer Simulation Technology (CST) is used to design and simulate the proposed antenna.The designed antenna radiator is in the form of a spike-shape. The four symmetrical monopoles are placed orthogonal to one other, resulting polarization diversity. The proposed spike-shaped MIMO antenna has size in the order of 40 mm × 40 mm × 1.6 mm  is used to study radiation patterns and scattering fields in perfect electric conductors. In perfect electric conductors, the antenna’s input impedance and radiation pattern are proportional to the total surface current density at the feeding point . The impn is zero at all points except the feed point of the conducting antenna. The overall phase of n.The dot product of The characteristics mode analysis (CMA) is used to develop the proposed antenna. The entire design process is carried out in four design stages performed in the proposed antenna design evaluation process. The total designing process of the suggested antenna depends on CMA metrics such as (1) Eigen values, (2) characteristic angle, and (3) modal significance. These three properties exist in each characteristic mode. In this current antenna design, only the modal significance parameter was considered. The Spike-shaped 4 × 4 MIMO antenna is analyzed and investigated in four stages: antenna0 (Ant#0), antenna1 (Ant#1), antenna2 (Ant#2), and antenna3 (Ant#3) . The CM The proposed antenna was developed using a CMA method. A series of sequential procedures are used throughout the designing process. The modal current distribution in Computer Simulation Technology (CST) is demonstrated using multi-layer solver. The evaluation process of antenna in step-by-step manner is depicted in Figure 2Modal significance is used to implement the proposed antenna in CMA design. Without using any excitation signal, the entire antenna design process is carried out in four evaluation stages from Ant#0 to Ant#3. The Ant#0 is made up of four spike-shaped radiators that are printed on FR-4 substrate. Each spike radiator has a radius of 4 mm with spike bubble radius of 1 mm. The Ant#0 has no conducting ground plane and is able to generate ten characteristic modes using the multilayer solver. Only five out of ten characteristic modes (CMs) contribute bandwidth and isolation. The modal significances of Ant#0 are shown in The remaining modes, CM1, CM2, CM4, CM5, CM6, CM8, CM9, and CM10, not only covers UWB but also the X-band (8 GHz–12 GHz). This antenna is suitable for UWB (3.1–10.6 GHz) and X-band applications (8–12 GHz). 11). The designed antenna provides isolation of 26 dB. This isolation is good in MIMO antenna metrics. The experimental set up for measuring the isolation using VNA is shown in 21 values. The novel type of decoupling structure in the ground plane provides the excellent isolation.The proposed UWB-MIMO spike-shaped antenna has good MIMO metrics such as radiation characteristics and isolation. The CMA technique is used to design the spike-shaped antenna, resulting in improved isolation. From 3.2 GHz to 12.44 GHz, the impedance bandwidth is 9.24 GHz and it covers the bandwidth requirements for UWB and ITU bands. Four performance metrics exist in MIMO antenna systems, all of which should be acceptable ,29,30. TAnother important MIMO diversity parameter is diversity gain (DG). Generally, in MIMO antennas the DG is nearly 10 dB. Formula  illustraThe acceptable value of CCL is 0.4 bits/s/Hz. In this proposed design, CCL of 0.31 bits/s/Hz is achieved and depicted in th elements. The MEG of the intended antenna is −3.1 dB, as illustrated in The MEG can be expressed using Equation (8) (8)MEGi=The spike-shaped 4 × 4 UWB-MIMO antenna is designed to work in both UWB and X-band. This antenna has some better features than others previously reported in the literature ,25,31,3211) are below −10 dB, and isolation between both the radiating elements are far better than 26 dB. The radiation parameters include 89% radiation efficiency, 118.15% impedance bandwidth, and a 4.9 dB gain. ECC, DG, MEG, and CCL have diversity features of 0.0016, 9.962 dB, −3.1 dB, and 0.31 bits/s/Hz, accordingly. In a cutting-edge experiment, these findings are verified. According to its performance parameters, the developed antenna is suitable for wireless communication in the spectrum of UWB and X-band.The spike-shaped UWB-MIMO antenna operates in the 3.2 GHz–12.44 GHz frequency range, which includes the entire UWB (3.1 GHz–10.6 GHz) and X-band (8 GHz–12 GHz). Four spike-shaped circular patch radiators are placed orthogonally to achieve polarization diversity. The reflection coefficients (S


In this paper, a novel wideband end-fire antenna, based on a spoof surface plasmon polaritons (SSPP) transmission line, is proposed. Periodically modulated corrugated metal strips are used as a transmission line for quasi-TEM conversion in the microstrip line to the state of SSPP and the best impedance matching. Due to the strong confinement of the field in the SSPP waveguide and its high transmission performance, it has been used as a transmission line. The antenna consists of SSPP waveguides for the transmission line, a metal plate on the ground as the reflector of the antenna, a metal strip director, and two half-rings to realize the radiation, reaching a wide bandwidth in the range of 4.1 to 8.1 GHz. The simulation results show that this antenna achieves a gain of 6.5 dBi, a bandwidth of 65%, and an efficiency of 97% across a wide operating frequency band, from 4.1 to 8.1 GHz. The proposed end-fire antenna has been fabricated, and the measured results agree well with the simulated results. The end-fire antenna implemented on a dielectric layer also has the advantages of high efficiency, good directivity, high gain, a wide bandwidth, easy fabrication, and a compact size. The Yagi-Uda antenna has a driving element, a reflector, and one or more directors.With recent advancements in wireless communication technology, the demand for low-profile wideband antennas is increasing. Yagi-Yoda antennas can be used for long-distance wireless communications due to their advantages, such as high gains, high bandwidths, low profiles, end-fire radiation patterns, and ease of fabrication and integration with other microwave circuits3. Moreover, the driver and director was reported that printed dipoles positioned on one side of a substrate with a high dielectric permittivity4; while the reflector was a truncated ground plane positioned on the other side of the substrate. The feeding method was geometrically complex, and needed long transmission lines4. In addition, the use of a narrowband delay line in the construction of the wideband antenna restricts the antenna's bandwidth, and creates an unbalanced operating situation5. In one study of quasi-Yagi antenna, a wide bandwidth of 44% was obtained using a simple structure and feed6; however, the asymmetrical nature of the printed quasi-Yagi antenna made the radiation pattern bad6. Also, a quasi-Yagi antenna based on a microstrip-to-slit transmission structure was presented, and it reached a bandwidth of approximately 46%7; however, that antenna had a large ground, which increased the size of the antenna7.In the literature, different designs have been reported to increase the bandwidths of planar-printed quasi-Yagi antennas. For example, a quasi-Yagi antenna based on a microstrip transmission structure was presented; and while the measured bandwidth was 48%, the structure of that antenna was complex8, which had an impedance bandwidth of 3.6 to 11.6 GHz, and a compact size8. Moreover, in another report of loop quasi -Yagi antenna, a loop Yagi-Uda antenna was used to reduce the antenna size by using the half-loop technique; and a measured bandwidth of 43.3% was obtained9. In addition, a comparison was made between two loop Yagi antennas—one with a defective ground structure (DGS), and the other without10. The antenna without the DGS had a better bandwidth, front-to-back ratio (FBR) gain and directivity, but it was 52% bulkier10. Therefore, it is challenging to design antennas with both high bandwidths and simple structures.Furthermore, a compact planar printed quasi -Yagi antenna with a modified ground plane, based on a new structure of microstrip-to-slit transmission line, was proposed11. The SPPs generate both charge oscillations in the metal and evanescent electromagnetic waves in the dielectric, and despite the fact that SPs and SPPs have been found at optical frequencies, they are closely related to surface waves at radio and microwave frequencies13. In 2004, for instance, Pendry et al. demonstrated a comparable surface plasmonic phenomenon known as SSPPs at microwave frequencies14.One way to increase the bandwidth is to use SSPP structures in the design of the antennas, where SPPs are the hybrid excitations produced when the SPs are coupled with electromagnetic (EM) waves17. Recently, SSPPs have been widely used in optoelectronic technologies18, filters21, amplifiers22, antennas30, and antennas reconfigure34, due to their special properties. In one paper, the authors used a metamaterial structure to achieve end-fire radiation26; moreover, in another one, high-gain and wideband SSPPs with circular patch arrays were presented27.SSPPs are therefore good candidates for high-density integrated circuits and components at millimeter and terahertz frequencies, due to their strong field confinements at the wavelength scale, as well as their groundless structures30. Furthermore, a small aperture end-fire antenna based on odd-mode SSPPs was presente11; where SSPPs antennas were composed of dipole-unit cells, which reached bandwidths in the range of 4–8 GHz11. However, most of the structures presented in the literature are not suitable for practical applications, due to their high fabrication costs and complex structures. Therefore, designing an antenna with a small size and high bandwidth is challenging.In one of the recent reports of SSPP, proposed a method for designing common aperture antennas based on (SSPPs) excitations, which are similar to microwave to (SPPs) in the optical frequency region. In this paper, a common aperture antenna has been designed based on both odd and even SSPP modes27. It is shown, here, that the bandwidth, efficiency, and gain of our proposed antenna are approximately 65%, 97%, and 6.6 dBi, at the designed frequency of 6 GHz, respectively.In this paper, an SSPP-based wideband antenna with high gains is proposed, where this SSPP waveguide is made up of a single ultra-thin, corrugated metal strip and a dielectric substrate layer. Here, the SSPP is used to transmit energy, while to realize the energy radiation, a Yagi loop has been added in the end-fire direction. In comparison to other transmission lines, this SSPP-fed antenna can simultaneously preserve the signal’s integrity, and it also incurs low conduction losses. To improve the directivity of the SSPP antenna, one director is also loaded at the end of the antenna. Our proposed antenna is simple, and has a better bandwidth, gain, and efficiency compared to other antennas presented in the literatureThe rest of the paper is organized as follows: Section “The configuration of our proposed wideband end-fire antenna, based on SSPP, is shown in Fig. mentclass2pt{minimThe designed antenna consists of five parts. A metal ground sheet is used on the bottom layer of the substrate, to improve the directionality, reduce the losses, and enhance the FBR of the antenna. Two slots created in front of the ground plane are then used as the antenna’s reflectors, which reflect the energy transmitted back, and this also improves the antenna’s radiation. Moreover, a pair of SSPP waveguides have been used as a transmission line to excite the end-fire antenna. This is due to its advantages, including high-field confinement, low losses, controllable dispersion, and slow waves. In addition, the tapered SSPP line provides superior impedance matching to open space, thereby suppressing any end-to-end reflections.35. In this paper, the gradient corrugated grooves structure is used to match the SSPP transmission line.Different methods are used in the literature for matching the SSPP transmission line. One of these methods is using the characteristic impedance matching method26, but we have extended this concept to excite end-fire loop Yagi antennas. Thus, our proposed design increases the bandwidth and efficiency, by preserving the same dimensions while also having a simpler structure compared to26. To achieve impedance matching, corrugated grooves have been used between the input microstrip line and the SSPP waveguide, where the parameters of the corrugated metal strips are given in Table It is worth noting that the concept of an SSPP waveguide was used in referenceAs can be seen in Fig. 36, meaning all of its parts contribute to the increase in radiation. According to the HWC for the end-fire radiation, the appropriate phase constant for the guided waves should be slightly larger than in the air. Yet, on the other hand, the dispersion curve needs to be under the airline, and also within a short distance. So, the dispersion curve of the unit cell has been simulated in CST software, as illustrated in Fig. It is worth noting that the proposed antenna is based on the Hanson-Woodyard condition (HWC)Therefore, the effect of the groove depth on the impedance matching has been analyzed, to achieve an optimum value of h10 for wide bandwidths in the frequency range of 4–8 GHz. As shown in Fig. The number of unit cells of the groove of the SPP transmission line have also been optimized, where a schematic of quasi-Yagi antennas with four different unit cell numbers is shown in Fig. The results of the gain and efficiency of our proposed antenna are shown in Fig. It is worth highlighting that the use of SSPP transmission lines has been considered, here, due to their lower losses and impedance matchings. To better understand this, the results of the reflection coefficients of the structure, both with and without the SSPP, are shown in Fig. The distribution of the electric field for the frequency of 6 GHz with SSPP and without SSPP is plotted in Fig. To experimentally verify the proposed wideband end-fire antenna based on an SSPP, as shown in Fig. mentclass2pt{minimFigure In this paper, a simple structure based on SSPPs, with a wide bandwidth, and a high efficiency and gain has been presented. Since the electromagnetic waves are strongly confined around the metal surface of this structure, the proposed antenna has low loss. Table In this paper, a compact quasi-Yagi antenna, based on an SSPP, has been designed, simulated, and fabricated. An SSPP waveguide has been used due to their ability to provide field confinement, slow waves, and low losses for the transmission line, and to obtain wide bandwidths. A high efficiency of 97%, with a wide bandwidth of 65%, and a peak gain of 6.5 dB has been obtained with the proposed antenna simulated on a Rogers Ro4350B substrate. However, due to the unavailability of this antenna has been performed on an FR-4 substrate. The bandwidth, maximum gain, and efficiency obtained from the simulation and measurement with the FR-4 substrate were approximately 61%, 6.1 dB, and 77%, respectively. Thus, based on the advantages of the SSPP transmission line, our proposed antenna with a wide bandwidth, high efficiency, and compact size, shows great potential for use in wireless communications.


A multi-layer beam-scanning leaky wave antenna (LWA) for remote vital sign monitoring (RVSM) at 60 GHz using a single-tone continuous-wave (CW) Doppler radar has been developed in a typical dynamic environment. The antenna’s components are: a partially reflecting surface (PRS), high-impedance surfaces (HISs), and a plain dielectric slab. A dipole antenna works as a source together with these elements to produce a gain of 24 dBi, a frequency beam scanning range of  Conventional methods of monitoring vital signs, such as an electrocardiogram (ECG), pulse oximetry, and capnography, require sensors to be attached directly to the patient’s body, which is either uncomfortable for the patient or not possible under certain circumstances ,5,6,7,8.The main advantages of using millimetre-wave frequencies for RVSM are (i) higher detection sensitivity due to shorter signal wavelengths, (ii) smaller form factors for more compact devices, and (iii) the ability to transmit and receive signals more precisely without interference from the environment ,9. ConveThe 60 GHz frequency band (57–66 GHz) has attracted RVSM interest due to its licence-free nature and its widespread use for several different wireless services. Antennas that operate in this band are particularly challenging to design and fabricate. RVSM measurements require the focus of the antenna beams, as well as enough gain to compensate for path loss in order to achieve high precision. Moreover, antenna beam steering capability is necessary to continuously monitor health with a fixed position sensor over a long period of time, especially when the person may move in practical scenarios, such as during sleep, inside a room, during transport in an ambulance, and at work ,12,13. SRecent publications ,8,9 discLeaky wave antennas (LWAs) are a special family of antennas that are able to steer their beam with frequency . They prOur study presents a multi-layer LWA used for remote vital signs detection in a typical situation in which a patient is lying on the bed and has some random movements during sleep. The non-contact detection of vital signs can be achieved using antenna beam characteristics that provide high gain (58–66 GHz) and wide bandwidth performance. A measured antenna bandwidth of 8 GHz and a maximum gain of 24 dBi are measured across the operating band. The proposed antennas are tested experimentally to validate their predicted detection coverage. This antenna design measures the respiratory rate (BR) and the heart rate (HR) from a distance of up to 4 m from the body of the person at five different radiation angles between This work is an improvement from the antennas already published in ,17. The This paper is organised as follows: The antenna design is described in P is selected for the HIS and PRS unit cells. The reflectance properties of PRS and HIS unit cells are determined at normal incidence. More details about this analysis technique and the relevant equations can be found in [To analyse the proposed design, we begin by applying ray optics in order to measure multiple reflections between partial reflective surfaces (PRSs) and high impedance surfaces (HISs) on the ground plane ,19. The found in ,21,22.Both PRSs and HISs are metasurface arrays made up of square patch elements and square ring elements, respectively, with a ground plane etched on a planar printed circuit board. The design process of the PRS and HIS unit cells is depicted in  antenna a. The to=1.57 mm b uses a TM along the x and y axis for both unit cell simulations. Open boundaries were established along the incident wave’s Periodic boundary conditions were utilised in CST Microwave Studioailed in  to extraIn this section, a printed dipole antenna (PDA) is designed, which will serve as a primary feeding for the complete leaky wave antenna structure. A design model of this feeding dipole antenna can be seen in xy plane . Figure The full proposed multi-layer LWA design is shown in d dipole a is posid dipole b. A 3D vP” can be predicted as:The height of the cavity between the PRS and the HIS ground plane, which is represented by the symbol duced in , the poiE-plane at various frequencies within the operational range, and The phase reflection coefficients of PRS and HIS as a function of frequencies are used to demonstrate LWA beam scanning. These coefficients are derived using Equation , which pfound in . The farTo prove the design concept, three additional variations of our proposed multi-layer LWA were considered, which have been demonstrated through simulations in CST Microwave Studio. Our proposed multi-layer LWA consists of a HIS-PRS cavity and a dielectric on top as in . Three vThe d in S11 a is wellThe entire antenna design, including the coaxial feed, is encased in a low-loss polytetrafluoroethylene (PTFE) so that it can be placed in the most realistic environment possible and measurements can be made. An external connection is made using a 1.85 mm flanged launcher and a GB185 glass bead. The antenna layers were fabricated using a typical low-cost PCB fabrication method. The measurement configuration for the multi-layer LWA’s The measurement of the antenna was carried out in an anechoic chamber. The gain-comparison method was used to determine the realised gain of our proposed multi-layer, leaky wave antenna. This method calls for two antennas: one that serves as a receive antenna and has a known gain  and another that serves as a transmitter antenna whose gain is unknown (our proposed multi-layer LWA). The antennas were placed at a far-field distance of 1 m. The antenna gain was next calculated using Friis’ equation. S-parameter stays lower than −10 dB. In addition, the beam scanning ranges of each of the tested antennas are the same, with a scanning range of A rotatable base connected to the computer for data acquisition was used to measure the patterns of antenna radiation. The antenna is rotated through 0°, 10°, 20°, 30°, up to 360° and then returned to the starting position after calibrating the VNA to 0 dB insertion loss. The information is then gathered and kept on file to forecast the signal strength. E-plane and H-plane radiation patterns at 62.5 GHz are presented in E-plane radiation patterns at different frequencies is given in To validate the design concept, the measured far-field radiation patterns of the co-polar and cross-polar in the Both the measured and simulated efficiency of antennas A and B remained at approximately e 60 GHz b. For thIn this experiment, we focus only on heart rate and respiration rate measured on the chest. Our proposed leaky wave antenna with a beam scanning range of d), reflected signals from the environment and the subject, and residual phase noise. The The proposed RVSM process is shown in a block diagram in ain (FD) : 3)R(t)R(t)(2)R, and the findings are updated after a shorter time interval  . High-reIn this experiment, the target is 2 m away from the transmitter and the receiver antennas at The estimated BR signal amplitude is seen in the first peak at a frequency of about 19 (1/min), while the predicted HR signal amplitude is seen in the second peak at a frequency of about 74 (1/min). At angles of  12° see . It is aThe HR peak is still visible at 66 GHz, and it is greater than the surrounding noise at 30° See e. It cane object .A hospital blood pressure monitor that also records patients’ heart rates on a daily basis was used to validate the results on the same individual, as shown in In order to demonstrate the antennas’ range at a radial distance of 4 m, the experiment was repeated multiple times with the same subject, who sat at varying angles away from the antennas each time and had their vital signs recorded. The purpose of this was to validate the results and compare them to those obtained from a range of 2 m. The other experimental parameters are kept as previously described. Three RVSM measurements were taken at the best frequencies and beam angles for maximum detect ability, according to The proposed multi-layer LWA design with Doppler radar technology is well-suited for use as a non-contact wireless sensor for health monitoring. As the experiment has shown, the sensor can detect both breathing rate (BR) and heart rate (HR) accurately and reliably at a radial distance of up to 4 metres. The non-contact nature of the sensor means that it can be used in a variety of scenarios, including when a patient is travelling in an ambulance or when they are sleeping or in occupational settings, where workers’ health can be monitored without the need for physical contact see . Additiopared to , which ipared to . HoweverThe proposed multi-layer leaky wave antenna is a highly promising technology due to its numerous advantages. It operates in the 60 GHz band, which is ideal for applications requiring high data rates, low interference, and short-high-range communication. Additionally, the antenna has a high gain and enough beam-steering range for RVSM  applications, which is essential for accurately tracking the movements of a person’s body. It is manufactured using low-cost PCB (Printed Circuit Board) fabrication methods, which makes it a highly attractive option for widespread adoption. Additionally, the design has been tested and characterised through two prototypes, and good agreement has been observed between the simulations and measurements. The experiment conducted to test the antennas for respiration and heartbeat detection showed that the proposed antenna is highly effective. The antenna was tested at radial distances of 2–4 m and angular positions of


This work presents an optically transparent and flexible MIMO antenna that features two square patch elements placed in close proximity, aiming to meet the demands of compactness, flexibility, optical transparency, and visual appeal for IoT applications and future 5G wireless communication. The design includes a simple offset fed configuration to achieve the required isolation and impedance matching. It simplifies the process of creating closely spaced transparent MIMO antenna configurations. By optimizing and analyzing this structure, the antenna achieves better isolation and diversity gain performance, even when the patch elements are positioned very close to each other. To achieve optical transparency and flexibility, the antenna uses thin polyethylene terephthalate (PET) material as a substrate, which is a thermoplastic polymer resin from the polyester family. The wired metal mesh parameters for conducting parts of the MIMO antenna and offset position of the feed are carefully optimized to achieve required optical transparency, isolation, impedance matching and radiation performance without any complex decoupling or impedance matching network. Transparent and flexible MIMO antennas serve as a key element to maintain compatibility with devices of varying shapes and sizes while minimizing their visual impact. The optical transparency may help to increase visual aesthetics of 5G-enabled base stations, smartphones, tablets and wide range of IoT devices, from environmental sensors to home automation devices. While the flexibility of the antenna design might make it suitable for integration into flexible communication modules or boards that can be adapted to various types of IoT sensor nodes and devices with different form factor.Multi-input-multi-output (MIMO) antenna systems are used to enhance spectral efficiency and channel capacities of 5G and IoT infrastructure3 for next-generation compact, transparent, and flexible 5G systems. However, a transparent and flexible MIMO antenna configuration is challenging to realize due to complex isolation techniques, such as vias and decoupling structures, which are not ideal for maintaining transparency and low-cost manufacturing processes5.These can also help to install more antennas in close proximity to enhance signal strength and data speeds6 ,7 despite having good flexibility. Furthermore, IZTO and ITO materials are not suitable for future transparent and flexible applications due to their brittleness and high levels of loss when bent, as referenced in8. Roll-to-roll production of monolayer graphene has been successfully demonstrated, boasting high transparency, electron mobility, and scalability. However, the transfer of large, high-quality graphene films remains a significant challenge, as mentioned in9 and10.During the past few years, State-of-the-art flexible and transparent MIMO antennas have been developed using advanced nanotechnology. Researchers have explored various conductive and optically transparent materials, including AgHTs (silver-coated polymers), IZTO (indium zinc thin oxide), ITO (indium tin oxide), and graphene, for their potential use in the development of transparent and flexible antennas for wearable applications. However, AgHTs suffer from high surface resistance and low radiation efficiency for antennas11, and all-solid-state supercapacitors12. However, there are only a few reported instances of stiff antennas using MMs as conductive electrodes in RF devices. There were reports of low efficiency and poor transparency in these MM antennas, as indicated in13 and14.The use of metallic mesh (MM) has been viewed as a promising technique for transparent and flexible applications owing to its superior transparency, conductivity, and malleability. The full potential of MM structures in contemporary radio frequency devices has not been completely investigated yet. In general, micro-structured MM films have found applications in solar cells, electromagnetic interference shielding15, transparent and flexible antennas are proposed, which offer good performance through the use of metallic mesh with embedded Nickel as conductive electrodes using a selective electrode position process in combination with inverted film-processing on polyethylene terephthalate (PET) substrates. A 4-element transparent flexible MIMO is presented for sub-6 GHz 5G applications in16. The researchers conducted parametric studies on antenna resonator geometry to ensure minimum fabrication errors, while maintaining high isolation levels across wide operation bandwidth. However, it is realized by carefully designing the connected ground plane lines and introducing complex slots in the substrate. Thus, developing flexible antennas with exceptional transparency and optimum efficiency is a challenging task. So, there is still need to achieve a simple and suitable solution for a transparent and flexible MIMO antenna configuration without using any decoupling structure between antenna elements, a defected ground structure or vias.InIn this study, a simple, transparent and flexible MIMO antenna with two close proximity patch antenna elements is proposed to achieve higher isolation without using any decoupling structure or complex matching network. The frequency band of the proposed MIMO antenna is centered at 4.5 GHz, which falls within the frequency range considered suitable for 5G New Radio (NR) sub-6 GHz band and IoT applications. An optically transparent square lattice of wired metal mesh is printed on a flexible transparent polyethylene terephthalate (PET) substrate to achieve transparency. A simple offset fed configuration is used to achieve higher isolation and impedance matching for the closely spaced patch antenna elements. The proposed transparent and flexible MIMO antenna is analyzed and optimized through parametric analysis and surface current distribution analysis of the square lattice of the wired metal mesh. The desired optical transparency, impedance matching, isolation, radiation performance, and diversity gain performance with flexibility are achieved and these are also verified through measurement of the fabricated prototype.To evaluate the effectiveness of a transparent MIMO antenna utilizing a wired metal mesh, a MIMO antenna operating in the 4.5 GHz band is chosen, with two patch elements located in close proximity to each other.Figure No decoupling or matching network is used to achieve isolation or impedance matching respectively, for the MIMO antenna configuration. Instead, a simple offset edge fed is used for excitation of both patch antenna elements. This simple asymmetric edge fed configuration is also suitable for development of large antenna arrays. The wired metal mesh square lattice is used in the proposed antenna to achieve approximately 83% optical transparency, with size of the square lattice denoted as mentclass2pt{minimFigure Generally, when two antennas are placed in close proximity to each other, these tend to couple or interact with each other, which can lead to poor isolation between the antennas. To overcome this issue, some decoupling structures and techniques are usually employed in literature to achieve better isolation. However, a decoupling structure between antenna elements may cause larger separation between antennas and may also increase power loss, which may limit compactness of MIMO antenna and may also reduce radiation efficiency.In this work, the offset edge fed plays a crucial role in achieving higher isolation between the patch antenna elements and good impedance matching.The fed point for both of the antennas is intentionally shifted or offset from the center of the antenna elements. By optimizing the offset position of the edge fed, higher isolation between the closely spaced patch elements as well as required impedance matching for each antenna and feed line can be achieved.A center fed conventional MIMO antenna configuration with same geometrical parameters and spacing between antenna elements as those of the proposed MIMO antenna is also analyzed to verify reason for enhanced isolation due to offset fed configuration and shown in Fig. The offset feding is used to control current distribution on each antenna element of MIMO antenna, thereby improving isolation without using any decoupling structure between antenna elements. Figure It can be seen that the offset fed configuration causes phased current distribution on each antenna element and the current distribution is asymmetric and little more concentrated towards offset fed point of antenna-1, and there is almost no current is seen on antenna-2. Similar but mirror phased current distribution is also observed on other antenna element when antenna-2 is excited, which is not shown here for brevity. The asymmetric current distribution in one antenna phased in opposite direction to that of the other antenna, which reduces the mutual coupling between closely spaced antenna elements. The degree of decoupling achieved through offset feeding depends on various parameters, including the offset distance, antenna geometry, and frequency of operation.Achieving optimal performance for multiple parameters simultaneously is a formidable task. Notably, there is still no significant work available in literature that is simultaneously focused to achieve desired optical transparency and flexibility for a MIMO antenna with closely spaced antenna elements. Optimization to achieve required transparency, flexibility, resonance frequency, impedance matching, isolation and radiation performance simultaneously, is a distinct and complex challenge because these performance metrics varies with the variation of the parameters of the square lattice of the wired metal mesh to achieve desired transparency as well.In Fig. entclass1pt{minima13:The definition of optical transparency (O.T) for square lattice of a wired metal mesh is given inAltering The The w = 0.2 mm, are selected for the square lattice to achieve optimum optical transparency of 83%, return loss of 22 dB, and isolation of 19 dB at 4.5 GHz. These optimized values strike a balance between transparency, resonance frequency, impedance matching and isolation for transparent and flexible MIMO antenna with closely spaced antenna elements.After this thorough analysis, we collectively determine the optimized values of The proposed transparent and flexible MIMO antenna has also been manufactured and tested to verify its simulated performance. Figure In addition to the previous analysis, the proposed transparent MIMO antenna is evaluated for its performance using measures such as radiation pattern, Envelope Correlation Coefficient (ECC), Channel Capacity Loss (CCL), Total Active Reflection Coefficient, and Mean Effective Gain (MEG).Figure The transparent patch antenna element demonstrates good radiation performance at 4.5 GHz, with a gain of 3.2 dB, a radiation efficiency of 61%, and total efficiency of 63%. This is quite impressive for a transparent and flexible antenna element. The 17. ECC performance of a MIMO antenna can be calculated through scattering parameters18, or through far-field patterns19. The following Equation (1919.17.Envelope Correlation Coefficient (ECC) is utilized to assess the effect of coupling between antenna elements, measuring the correlation between them in a MIMO diversity system. Higher diversity gain performance is associated with a lower ECC value that should be less than 0.5Equation  is used wing Eq.  mentione17. According to Eq. Figure while Comparative analysis of the proposed work with previous recent similar works is also presented in Table A transparent and flexible MIMO antenna design is presented in this study, featuring patch elements in close proximity, which are developed with a wired metal mesh having a square lattice to achieve optical transparency. The substrate material used is a 0.1 mm thick transparent and flexible PET material. To achieve isolation, an offset fed configuration is used, which delivers more than 19 dB measured isolation in the desired frequency band. The proposed MIMO antenna is optimized and analyzed based on the square lattice wired metal mesh parameters and the offset position of the edge fed. The proposed MIMO antenna has achieved 83% transparency and operates at 4.5 GHz. The performance of the proposed transparent and flexible MIMO antenna is evaluated based on parameters such as gain, radiation efficiency, total efficiency, envelope correlation coefficient, diversity gain, TARC, and mean effective gain. The results of the fabricated prototype measurement confirm the suitability of the proposed transparent and flexible MIMO antenna for compact applications requiring optical transparency and visual aesthetic with flexibility. The presented technique could play a crucial role in developing future smart wireless transparent and flexible devices.


A self-isolated multiple-input-multiple-output (MIMO) antenna in a compact shared ground structure is proposed for 5G systems. The proposed MIMO antenna consists of customized M-pattern, closely coupled members. It benefits to attain good isolation of the targeted bandwidth transversely without additional de-coupling structures. It is discovered that the arm of M-pattern antenna members can cancel out the coupling on the system and achieve sound isolation among antenna members. A relevant matching circuit model is discussed to show how the suggested theory works in principle. The mixed couplings among antenna members neutralize by modifying the antenna shape with the help of electric and magnetic coupling and surface currents. The proposed self-isolated 2-member MIMO antenna demonstrates sound isolation superior to 15 dB transversely in the frequency bands dedicated to 5G NR: n48/n78 and long-term evolution (LTE) band 42/43/48/52 (3.2–3.98 GHz). Moreover, the proposed 2-member design structure has a scalability advantage. It is extended into an 8-members structure, where a pair of antennas located at each side of the frame offers orthogonality. The proposed 8-members M-pattern MIMO is validated using fabricated and simulated measurements. The investigational results show that the 8-members MIMO (M-shaped) system is effective in higher order MIMO antenna design and offers more than 20 dB isolation transversely in the frequency band 5G NR n48 and LTE band 42/43/52(3.29–3.66 GHz). Particular of the concerning issues is the short isolation among the closely coupled antennas that are utilized in a compact device, which weakens the MIMO systems performance3. So, it is precarious and essential to discover self-deprecating and dynamic isolation methods for the MIMO system.MIMO technology is extensively employed in wireless communication systems to address the growing need for better service quality, lower latency, and higher data rates. Several antennas lay on in the system have encountered new issues in fulfilling the overhead demands14. The narrow open-ended slots and space diversity3, DGS in the ground plane4,7, the vertical ground strip with EBG structure in the top plane orthogonally arranged patch antenna5, the shorting stub6, the complementary-split‐ring-array (CSRA)8, loading split-ring-resonators (SRR)11, complementary slot coupled SRR12, the planar EBG structure13, 2nd order de-coupling circuit14 are applied in the MIMO antenna, which in turn leads to isolation improvement. Phase-gradient metasurfaces (PGMs) are used to deflect an antenna beam in a preferred direction15. The hybrid dispersion-engineered metamirrors for dispersion manipulation of the antenna are utilized16. Nonetheless, most overhead works increase isolation and change the antenna properties with the help of supplementary design, which could change the characteristics of the antenna. Nevertheless, these techniques upsurge the intricacy of the design pattern with a rising number of members. Although the techniques revealed above show appealing traits, they experience issues and complexities generated with the aid of supplementary systems.Numerous approaches in the previous works of MIMO antennas to increase the isolation have neutralization line, defected ground structure (DGS), metamaterial, and Electromagnetic Band Gap (EBG)25. There are techniques that works with common coupled grounding branch17, common-mode(CM) and differential-mode(DM) tunning18, using ground plane weak-field19, gap-coupled loop antennas using asymmetrically mirrored arrangement20, L-pattern branch operates as a radiating and de-coupling element21, inverted U-pattern member22, using closely paired23, couplings between a pair of coupled antennas are canceled out by proper adjustment25. The shorting legs in the PIFA pair6 are utilized to attain self-isolation in the MIMO antenna. However, one of the above works primarily employs an opposed location to de-couple the antenna members. Moreover, one paper offered a weak field concept, and one more displayed signaling to tune using CM and DM concepts to de-couple the antenna members. Looking at the self-isolation MIMO antenna system's ostensible benefits is necessary.Various self-isolation-based approaches are offered in literatureThe mixed-coupling effect using the close coupling technique is introduced and managed without additional de-coupling circuits with the neighboring members. The magnetic and electric coupling combination agreements surface current cancellation and provide self-isolation.The central concept is to equilibrium the coupling between the antenna members employing stretched arms adjusting mixed-coupling among the elements.The presented structure is compressed, likened to earlier papers because the system's tightly coupling designed structure attains the self-isolation result. The monopoles of the antenna are placed close together to offer sufficient capacitive and inductive coupling to the other element.The proposed MIMO antenna members are easily extended to more than a 2-member array due to the compactness and easily integrable of the proposed design structure.The flexible integrability of waveguide and coaxial feed is achieved easily due to compactness and self-isolation scenarios as presented in the design.g at 3.8 GHz).The MIMO antenna self-isolation characteristics deal with isolations better than 16 dB for the 2-members structure and 20 dB for the 8-members design throughout the operating band. The shortest distance among antenna members is about 0.2 mm  antenna can offer the 3.5 GHz band (3.2–3.98 GHz), and the 8-member MIMO (M-shaped) antenna can cover the 3.5 GHz band (3.29–3.66 GHz), which is directed for the approaching 5G band for sub-6 GHz range. MIMO antenna pair with a common ground to develop MIMO antenna integration levels and house the upcoming 5G systems. The mobile frame-based eight members MIMO antenna example is presented with user hand effect analysis.The self-isolated closely coupled MIMO (M-shaped) antenna member is represented in Fig. 4 dB isolation of coupled antenna members offered at the center frequency. The length I5 of the T-pattern configuration adjusts the center frequency, as presented in Fig. In the initial configuration, a T-pattern member is employed in the MIMO antenna projected as AntI, as shown in the inset of Fig. 0.2 mm (0.005λg at 3.8 GHz). The customized arm of the presented structure provides sound isolation among the antenna members with the help of ground cuts I6, and I7.In the next evolution, the T-pattern structure is customized as AntII, as shown in the inset of Fig. The passive matching circuit is used to comprehensively analyze the projected self-isolated MIMO antenna. The MIMO antenna suffers coupling owing to capacitive and inductive influences between the members. The inductive and capacitive effects occur due to the adjustment of the structure. The maximum electric field between the members offers capacitive coupling, and magnetic fields at the edge of the customized design offer inductive coupling. This effect can be manipulated deliberately and abolish the coupling between the pair of members with the same magnitude with the help of surface current. The close-coupled arm of the antenna member produces surface currents to the equilibrium of the coupling introduced by the nearby antenna member. The following section describes the self-isolated designed MIMO (M-shaped) antenna matching passive circuit to determine the appropriate configuration.28. The antenna members are represented by members L1, C1, L2, and R2. The coupling among the antenna members is presented inside the dotted red box as mutual admittance Y21,Cou, consisting of L3, C3, LM, and CM. The coupled antenna members consist of two coupling types (inductive and capacitive). The mixed inductive and capacitive coupling is represented in parallel circuit members. To cancel these couplings, conscious tuning is required. The coupling among antenna members is defined stated as CM and LM for capacitive and inductive coupling matching. In this instance, the mutual couplings CM and LM are capacitors and inductors among antenna members. The impact of the coupling influence can be analyzed using an enclosed circuit inside the red box. The mutual coupling influence of the proposed MIMO (M-shaped) antenna is confined in Fig. 21,Cou the coupling admittance needs to be modified; this helps to strengthen Y21 total mutual admittance. The Y21,Cou can be signified as:The matching passive circuit is used to explain the concept of the projected MIMO antenna's structure, presented in Fig. L3, CM, and LM adjusting can cancel out Y21,Cou at a particular frequency. The circuit component values are estimated with the help of the Keysight ADS Simulator, and circuit values are listed in Table R2 are Ω, L1, L2, L3, LM, are in nH, C1, C3, CM, are in pF, and Y21,Cou in mho. The Y21,Cou mutual admittance at the middle frequency of AntI, and AntII is -j4.057 and –j0.908, respectively. This displays good cancellation between the self-isolated MIMO antenna. The projected MIMO antenna 3D model and the phase plots and S-parameters simulated results of the matching passive circuit are shown in Figs. Here ω represents radian frequency. The parameters I4 differs from the self-isolated MIMO (M-shaped) antenna's central frequency. It is explicable because I4 is the size of the radiating arm of the AntII. It is also noticeable that the isolation of the members is also tuned with it as well. The arm's width also impacts the S-parameters of the MIMO (M-shaped) antenna described in Fig. I6 and I7 also helps to adjust the isolation of the MIMO antenna. The parametric variation due to I6 and I7 are illustrated in Fig. It is realized from Fig. I4, W4, I5, and I3 dimensions. The closely coupled arms of the MIMO members help improve the structure's isolation by coupled surface current neutralization, as demonstrated in Fig. The customized M-pattern MIMO system performance can be improved by appropriately choosing the The projected customized M-pattern MIMO antenna radiates at 3.8 GHz with more than 25 dB isolation. The introduced MIMO antenna radiates from 3.5-4 GHz with an isolation of more than 15 dB. After the evolution of the customized MIMO (M-shaped) antenna, the improved dimensions are shown in Fig. Although the antenna members are tightly connected, this mixed coupled procedure enables the self-isolated characteristic of the offered MIMO antennas. An 8-member MIMO (M-shaped) antenna system is additionally validated for 5G systems, where the experimental results authenticate the viability of the projected technique.As discussed in the last segment, the 2-member MIMO antenna presents self-isolating properties with excellent isolation in compact dimension isolation using mixed-coupling. These antenna member pairs can be extended to each side of the frame using a similar structure or the customized design shown in this section.140 mm × 70 mm × 1.6 mm, while the ground size is similar to the dimension of the system with defects. A pair of closely coupled MIMO antennas are designed in the proposed system located at each side of the frame. The pair's orthogonality can help to improve the isolation among the other pairs of self-isolated MIMO antennas.The closely coupled compact self-isolated 8-member MIMO antenna member is described in Fig. 0.0047λg (0.2 mm) at 3.45 GHz. The Ant1 radiates from 3.26–3.6 GHz, and the Ant5 radiates from 3.26–3.63 GHz. The minimum isolation in the projected 8-member MIMO antenna is 17 dB at 3.45 GHz.The MIMO antenna comprises dual pairs of similar M-pattern antenna members , as shown in Figs. The radiating frequency and isolation of the MIMO (M-shaped) antenna adjust along with the dimension of the M-pattern arm and ground defects; some parametric outcomes associated with the design process are shown in Fig. The closely-coupled improved MIMO antenna is manufactured with the help of the S63 LPKF Machine, shown in Fig. S-parameter results of the optimized closely coupled M-pattern MIMO antenna are matched with the simulated in Fig. 29. The assessed outcomes validate that the projected MIMO (M-shaped) antenna system has high efficiency and excellent isolation.The polar plots of recommended 2-member closely coupled customized M-pattern MIMO antenna are assessed at 3.65 GHz, as demonstrated in Fig. 30. The 3D E-fields measured for assessing ECC, as presented in the formula in 2. The projected design ECC is demonstrated in Fig. 31.ith antenna e-fields, when another antenna in the design is fitted with the matched load of 50Ω.Consequently, the ECC of the self-isolated 2-member MIMO (M-shaped) antenna is evaluatedCloss). The maximum information rate that may reliably be transmitted through the channel characterizes the channel's capacity, matter-to-channel properties, and the radiation from the antenna. The formula shown as 3 can be used to evaluate the MIMO antenna Closs., as explained in32. Its estimate is below 0.0125 b/s/Hz, as displayed in Fig. i, j = 1 or 2.The supplementary important diversity characteristic is the channel capacity loss (The 8-member tightly coupled MIMO (M-shaped) antenna is manufactured utilizing the LPKF machine, as displayed in Fig. 18. The system dimension is 143.2 mm × 73.2 mm × 1.6 mm, while the ground plane size is 140 mm × 70 mm . The pairs of antenna members are arranged similarly to an 8-member system in mobile frame, with a depth of 1.6 mm and a height of 10 mm. The frame of the antenna considered during design is an FR4 substrate. The S-parameters simulated results of the mobile frame-based MIMO (M-shaped) antenna are presented in Fig. Figure The user’s hand effects are analyzed foresight member mobile frame-based MIMO antenna. CST Microwave Studio, the single-handhold arrangement, examines and simulates the proposed antenna. The simulation-based user hands model's front and back views are presented in Fig. 22, the projected antenna introduces a good reflection coefficient and isolation without a physical link. Other works17,21 utilize the current elimination or reorganization method to enhance the isolation with lesser bandwidth assessed to the projected design. In work19, weak field and space diversity methods achieve isolation for limited bandwidth. The asymmetrical arrangement achieves low impedance bandwidth and low isolation with better efficiency20. Coupling is terminated by nearby coupling and coupling elimination method for restricted bandwidth24.Above that, the self-isolation antenna design is compared to nearly earlier work in Table In this article, a compact and straightforward self-isolated MIMO (M-shaped) antenna is offered by putting together M-pattern members nearby. It has been established that high isolation is attained due to the closely coupled members canceling out the coupled members' surface currents. The 2-member MIMO antenna radiated a 3.2–3.98 GHz frequency range with 16 dB isolation, and in the frequency range of 3.5–3.82 GHz, more than 20 dB isolation occurred. The diversity characteristics ECC of this antenna are healthier than 0.175. The suggested M-shaped self-isolation MIMO design method can be utilized for a larger array size in MIMO. The 5G application 8-member MIMO antennas are explored in simulation and measurement to validate that. The 8-member MIMO antenna offers 20 dB isolation in the 3.29–3.66 GHz frequency band. The antennas can provide good return loss and isolation with decent diversity characteristics. With the rewards of self-isolation, common ground, wide bandwidth, simple structure, and high efficiency, the projected 5G smartphone arrangement of MIMO antenna design exhibit a promising future.


A triple-band substrate integrated waveguide (SIW) with dielectric resonator antenna (DRA) for fourth-generation (4G) and fifth-generation (5G) applications is proposed and analyzed in this paper. Loading SIW with DRA allows for a wide bandwidth, low losses, and fabrication ease. The proposed antenna can transmit and receive data independently by covering LTE Band 3 at 1.8 GHz, LTE Band 8 at 2.6 GHz, and 5G n77 at 3.7 GHz. A U-shaped cut is applied to achieve the targeted multi-resonance frequencies. The antenna obtains high bandwidths of up to 19.50% with 4.9 dBi gain and 81.0% efficiency at 1.8 GHz, 6.58% bandwidth with 4.4 dBi and 72.7% efficiency at 2.6 GHz, and 8.21% bandwidth with 6.7 dBi and 73.5% efficiency at 3.7 GHz. The simulated and measured results agree well. The proposed antenna is feasible for 4G and 5G applications. An antenna with a compact size and multiple bands is preferred in this modern era of wireless communication ,2. SubstDielectric Resonator Antennas (DRAs) are used extensively in the electromagnetic field in comparison to microstrip antennas because of their superior qualities such as higher radiation efficiency, higher gain, and greater bandwidth . SeveralIn this paper, a rectangular DRA excited by a U-shaped cut in SIW is presented for 4G and 5G applications. The U-shaped cut is -utilized to generate multi resonance frequencies. The simulation of the presented antenna is performed with the help of Ansoft HFSS version 2019. This article is structured as follows: The proposed antenna’s design specifications and the development of the antenna design are covered in A 3D model view of a single port SIW with a DRA is shown in It is excited in its respective mode by the U-shaped cut configuration, which is located in the top wall of SIW and operates in  and (6) :(5)wd=wεEquation (7) is further refined by including d/ameter d .(7)WsiwThe metalized via holes are designed to create SIW. The SIW consists of two rows of metallic vias of diameter, d and separated by a distance, p. To keep energy leakage to a minimal , the valTo avoid any bandgap effects in the operating bandwidth, the following equation should be calculated :(11)pλc<One end of the SIW is short-circuited to produce standing waves inside the SIW . A microSix different antenna configurations of the single-port SIW with a DRA are investigated to resonate at 1.8 GHz, 2.6 GHz, and 3.7 GHz to cover the sub-6 GHz frequency for the 5G wireless communication system while also being backwards compatible with 4G bands. The evolutions of the proposed antenna in top view and reflection coefficient are depicted in Based on The SIW with a transverse cut is then loaded with a DRA and placed in the SIW’s centre, and the antenna is denoted as ‘ANT A’ in As depicted in In y-axis need to be properly chosen as they affect the coupling of energy to the DRA element. Based on ‘ANT D’ presents the proposed antenna design as shown in The reflection coefficient between the U-shaped cut SIW antenna and the proposed antenna has been compared in In this work, the reflection coefficient of U-shaped cut is compared to Z-shaped, inverted Z-shaped, H-shaped, and inverted U-shaped is depicted in In this section, the influence of design parameters on the bandwidth response of the proposed antenna has been carried out. Parametric studies are conducted for optimal parameter selection. Firstly, the effects of U-shaped cut size and its position are studied to determine the best coupling to the DRA. x-axis, sx has a negligible effect on the bandwidth at the three frequency bands. The only difference is the shift of resonant frequency at lower frequency band when sx increases. When sx = 0 mm, the antenna resonates at 1.8 GHz sharply where the maximum electric field flows through y-axis, sy affects the reflection coefficients at the lower and upper resonant frequencies only. A decrease of sy shifts both resonance frequency of lower and higher frequency bands to lower. When the U-shaped cut is not too close to the microstrip transition where sy = −2 mm, it has the widest impedance bandwidth at 1.8 GHz and is resonating at 1.8 GHz and 3.7 GHz.Based on The length and width of the DRA, denoted as x1 and y1 in x-axis, x2 has a negligible effect on the reflection coefficient at the three frequency bands. However, a change in DRA position in y-axis, y2 affects the reflection coefficients at the middle resonant frequency. When y2 = −2 mm which is closer to the microstrip transition, it gives the best coupling of the energy for the U-shaped cut to the DRA element at 2.6 GHz. Based on The top and bottom views of the prototype of the single port SIW with a DRA are shown in All simulated radiation patterns point in broadside directions for all three frequencies except the E-plane radiation pattern at 2.6 GHz, which points at 45°, as shown in A multiband SIW with a single-element DRA is designed, fabricated, and tested. The U-shaped cut is used to enable multi-band operation at frequencies of 1.8 GHz, 2.6 GHz, and 3.7 GHz while providing a compact antenna size. The proposed antenna demonstrates a high bandwidth and a good gain at each frequency band. The simulated and measured results agree well. This research can be extended by applying Multiple-Input Multiple-Output (MIMO) technology by utilizing multiple antennas at the transmitting and receiving ends of the communication system  to comba


Beam-switching is one of the paramount focuses of 28 GHz millimeter-wave 5G devices. In this paper, a one-dimensional (1D) pattern reconfigurable leaky-wave antenna (LWA) was investigated and developed for wireless terminals. In order to provide a cost-effective solution, a uniform half-width LWA was used. The 1D beam-switching LWA was designed using three feed points at three different positions; by selecting the feeds, the direction of the beam can be switched. The antenna can switch the beam in three different directions along the antenna axis, such as backward, broadside, and forward. The 1D beam-switching antenna was fabricated, and because of the wide beamwidth, the measured radiation patterns can fill 128 Increasing demand for high data throughput and reliable mobile services has motivated the development of the 5G mobile communication standard ,9,10,11.Various antennas have been reported so far for 5G millimeter-wave communications. For example, a dual, linearly polarized antenna element with four substrate layers and three films for bonding was proposed in . Then, fications . The 5×5posed in . The gaiposed in , a squarposed in , a quasiposed in  for applguide in . Sixty pantennas ,40,41,42antennas ,44,45. LTo avoid the above-mentioned issues and enable a cost-effective solution for wireless terminal beam-switching antennas in the millimeter-wave frequency range, a low-cost yet effective solution is presented here. The losses associated with the switching used in this work are negligible. Firstly, a 1D beam-switching antenna was designed using a single radiating element with three feed points at three different positions. Two antennas were then placed at a right angle to achieve 2D beam switching. Both 1D and 2D beam-switching antennas were fabricated and measured. The 2D beam-switching antenna has continuous 3 dB coverage from The objective was to design a simple beam-switching antenna for 28 GHz 5G wireless terminals. It is known that a uniform leaky-wave antenna (LWA) scans the beam in the off-broadside region  of the microstrip line is 54.4 mm or 5.1As shown in tor at I a is termL = 5.1The antenna in ented in , is showL = 5.1The normalized x-z plane radiation patterns of the HW-MLWA with length  = 5.1λ0 a are shoirection . With a L = 19.2 mm or 1.8For millimeter-wave wireless devices, we need a shorter antenna that takes only a small area of the printed circuit board (PCB) since a wider beam is preferred. It is observed from the surface current distribution in L = 1.8The input reflection coefficient of the LWA  in addition to point T. For this purpose, in Step 3, the two loading vias are removed, and the corresponding reflection coefficient is shown in The absence of loading vias has a very negligible effect on the y-z plane radiation patterns within the frequency range of 27–30 GHz. Here, we compare the radiation patterns of the antenna in the frequency range of 27–30 GHz with loading vias see b and witThis section presents the design of a 1D beam-switching antenna for 28 GHz 5G wireless terminals, which can point to the beam in three different directions along the antenna axis.The properties of the previous two antenna designs are incorporated into a single antenna to achieve 1D beam switching using a single radiating element.The antenna is fed at three different locations, i.e., left feed (LF), right feed (RF), and center feed (CF), as shown in The two steps of evolution of the 1D pattern reconfigurable antenna are depicted in In Step 2, an arrangement for the center feed is added to the antenna together with a trapezoidal patch, which is aligned to the free edge GU, as shown in The effect of the CF on the HPBW of the antenna in x-z and y-z planes when LF/RF is ‘ON’ is presented in When a side feed LF/RF is ‘ON’, the center feed restricts energy traveling towards the other side of the antenna see a. TherefTo validate the 1D antenna concept, a prototype was fabricated and measured. It is shown in The measured S-parameters of the 1D pattern reconfigurable antenna are shown in The measured transmission coefficients  with dimensions of 120 mm × 70 mm is shown in From the S-parameter measurements, we found excellent agreement between the predicted and measured results and, hence, for better presentation, only measured results are shown here in The measured transmission coefficients  radiating antenna was designed for the 28 GHz millimeter-wave band. Following that, an antenna was designed for radiating in the broadside direction. Using the properties of both antennas, a novel single-element LWA with three ports was developed to achieve 1D beam switching. The 1D beam-switching antenna was prototyped and tested to prove the concept. Using the three feeds, the beam can be pointed at