## 3 Results and Discussion
### Phase-Amplitude Resonant Measurement Method
The load acquires the desired inductive character (\(f>f_{v}\)) when the temperature is above 250 \({}^{\circ}\)C.  
It is known from [62], that resonance frequency \(f_{\rm T}\), and resonance frequency \(f_{v}\) defined as follows (10):  
\[f_{\rm T}=\frac{1}{\sqrt{\rm C_{1}L_{1}}},\quad f_{v}=\frac{1}{\sqrt{(\rm C_{ 1}L_{1}L_{2})/(L_{1}+L_{2})}}. \tag{10}\]  
Figure 8: Simulated model of a serial-parallel resonant circuit of the LCL topology in the Multi-sim environment. Figure 9: Controlled section of the PFR of the inductor L\({}_{1}\) circuit. Using the PSM method when controlling a full-bridge invertor topology loaded with an LCL circuit and operating on the \(f_{v}\) frequency provides the moment when the current \(I\) of the inductor L\({}_{1}\) passes through zero with an active load (\(f=f_{v}\)) at the moment O, corresponding to half the phase shift \(\chi\) between the switching moments of the left and the right pair of transistors, that is, in the middle (O \(\equiv\chi/2\)) between the front edges of the control signals Q\({}_{1}\) and Q\({}_{2}\) (or Q\({}_{3}\) and Q\({}_{4}\)). A change in the temperature of the nozzle leads to the fact that the moment when the current \(I\) passes through zero is shifted relative to the moment O. Due to the active nature of the L\({}_{3}\)D\({}_{5}\)D\({}_{6}\)R\({}_{1}\) series circuit, the induced EMF in the L\({}_{3}\) coil and the voltage (useful signal U\({}_{\rm T}\)) on the limiter diode D\({}_{5}\)D\({}_{6}\) are in phase with the current \(I\). The useful signal U\({}_{\rm T}\) is amplified and shifted to the positive area by introducing a constant offset into the signal, and then subjected to analog-to-digital conversion. Changes in U\({}_{\rm T}\) at the input of the operational amplifier from \(-8.25\) to \(8.25\) mV are registered. The moments of activation of the analog-to-digital converter (ADC) are synchronized (Figure 10) with the control signals Q\({}_{1}\) and Q\({}_{2}\) (with the moments of switching transistors) using a common clock source K. Then, each period of the current \(I\) of the inductor the moment O \(\equiv\chi/2\) is determined, after which using the ADC, a pair of successive measurements is made with a time shift n before and after the moment O (measurement a at the moment O-n, measurement b at the moment O+n), then the constant offset of the useful signal is eliminated, after which the successive measurements a and b are summed. The sum of measurements a and b characterizes the phase shift \(\varphi\) between the current signal of the inductor circuit and the voltage signal at the output of the HF generator, as well as the signal U\({}_{\rm T}\) amplitude. The sum is interpreted in terms of the generalized parameter \(\Omega\), that is measured in millivolts. As in our previous paper [53], the oversampling and averaging method [63; 64] as well as the exponential moving average filter were used to improve the measurement accuracy. The soft switching of transistors, which is ensured when the developed device operates at a frequency \(f>f_{v}\), which is known from [54; 62; 65], also contributes to the transistor switching noise abatement. To convert the measured values \(\Omega\) into the nozzle temperature \(T\), there were used the regression model created in [53] for the amplitude method. This turned out to be possible due to the fact that information about the magnitude of the phase change can be obtained with two specific measurements of the signal amplitude and linear interpolation of the results. Calibration process of the temperature readings was performed in a closed-loop nozzle temperature control system at \(V_{ext}\) = 0 mm/s during several sequential experiment.