## 5 Antenna Processing and Testing  
The simulated antenna in Figure 3 was fabricated into a physical prototype. To address the wide slot dimensions and small waveguide diameter, the cavity material was changed  
Figure 5: Distribution of electric field on the surface of mirror symmetric slit with time: (**a**) 0; (**b**) T/4; (c) T/2; (d) 3T/4. Figure 6: Comparison of reflection coefficient between equidistant and non-equidistant. from an aluminum alloy to stainless steel to enhance its structural rigidity. Considering the harsh marine environment's impact on communication performance, a corrosion-resistant fiberglass-reinforced polymer (FRP) radome was integrated into the housing. This radome ensures excellent electromagnetic transparency while prolonging the antenna's operational lifespan. The radome covers both the slot radiation region and impedance matching section, with additional potting encapsulation applied to seal critical components. The housing structure employs riveted connections between the feeding port and impedance matching section, as detailed in Figure 7.  
The antenna was connected to a vector network analyzer to obtain the parameters \(|\)S111\(|\). The axial ratio and direction pattern of the antenna were tested in a microwave darkroom, and the simulation and measured data of the antenna in the 1.4-2.8 GHz band were obtained, as shown in Figure 8. Among them, the measured S11 of the antenna is roughly consistent with the simulation data, and the measured antenna data are shown in Figure 8a. The range of \(|\)S11\(|\)\(\leq-10\) dB is 1.59-1.84 GHz and 2.48-2.55 GHz, both of which include the upstream and downstream frequency bands of BeiDou-1. Figure 8: Antenna simulation \(|\)S11\(|\) data: **(a)** data diagram; **(b)** field diagram. Figure 7: Physical diagram of the antenna: **(a)** description of the structure of the object; **(b)** Side view of the antenna when the FRP cover is added; (c) comparison of the gap of the antenna; **(d)** feed port picture. Figure 9 shows the ratio of the measured gain to the axis of the antenna. It can be seen that the measured gain of the antenna is slightly lower than the simulation result. The maximum gain at 1.6 GHz is 4.1 dBi, and the measured axis ratio of the antenna at 1.60-1.62 GHz and 2.4-2.5 GHz is slightly equal to 3 dB, which is in high coincidence with the simulation result. Figures 10 and 11 show the left-right circular polarization gain of the antenna at 1.62 GHz and 2.49 GHz, respectively. It can be found that the measured right-right circular polarization gain of the antenna at the two frequency points has a higher coincidence degree with the simulated gain, and it has good omnidirectional radiation characteristics. The main reason for the actual field distribution being different from the simulation environment is the different materials used for the antenna feed end and the radiation end at the riveting joint. The second reason for the difference is the limitation of the dimensional accuracy of the antenna gap during processing. The dual-band operation of the antenna arises from the superposition of multiple design factors. During simulation, the frequency sweep was conducted over 1.4-2.8 GHz. When calculating the cascaded coaxial impedance using Equation (7), only the resonant frequencies at 1.62 GHz (L-band) and 2.49 GHz (S-band) were prioritized, while impedance matching across the intermediate frequency range was not rigorously addressed. Furthermore, structural discontinuities between the tapered line segments, cascaded coaxial sections, and the coaxial feed of the rear radiating unit induce uneven field distribution within the waveguide cavity [10]. This configuration suppresses the impact of higher-order modes (e.g., TE\({}_{21}\), TM\({}_{11}\)) on the fundamental TE\({}_{11}\) and TM\({}_{01}\) modes in the L-band. Finally, fine-tuning of the inter-element spacing was implemented to optimize both resonant frequencies, achieving wideband operation across the target bands.