# A High Power Factor LED Driver with Intrinsic Current Balancing Capability
## 4 Circuit Design and Experimental Results
### Experimental Results  
The control circuitry was composed of a microcontroller (dsPIC33 FJ16 GS504) and two power MOSFET drivers (TLP250). From Equation (20), the LED power can be controlled by regulating the switching frequency of the active switches. A current sensor (ACS712) was used to sense the current in one of the LED strings and send its output to the microcontroller. Then, the frequency was adjusted to achieve stable LED current. A prototype circuit with the component parameters, as shown in Table 2, was built and tested. The experimental results at rated power are shown in Figure 6a-g. Figure 6a shows the primary current (the lower waveform) and the secondary current (the upper waveform) in \(T_{1}\) measured near the peak input voltage. Since the turn ratio of \(T_{1}\) is 2, it can be seen that the peak value of the secondary current is twice that of the primary current. In addition, the current waveforms indicate that the flyback converter is operated at DCM, as expected. Figure 6b shows the voltage and current of the input line. Both of them are sinusoidal waveforms. As shown, the input current can follow the waveform of the line voltage and in phase with each other. It ensures a high power factor and low THDi. The measured power factor of 0.989 and THDi of 5.27% validate the excellent functionality of the flyback-typed PFC circuit. The drain-to-source voltage and current of \(S_{1}\) are shown in Figure 6c. By designing the series resonant converter to present inductive characteristics, the resonant current will lag the input voltage of the resonant converter. Therefore, as soon as \(S_{2}\) is turned off, the resonant current is converted to flow through the parasitic capacitor of \(S_{1}\). When the parasitic capacitor is discharged to near zero volts (0.7 V), the resonant current will flow through \(D_{s1}\) and then the voltage across \(S_{1}\) is clamped to almost zero volts and the ZVS operation of \(S_{1}\) is achieved, resulting in low switching losses. However, the shared active switch \(S_{2}\) cannot fulfill ZVS operation. The drain-to-source voltage and current of \(S_{2}\) are shown in Figure 6d. Since \(D_{2}\) prevents the parasitic capacitance of \(S_{2}\) from discharging to the resonant circuit, a spike current happens due to the short-circuit discharge of the parasitic capacitance as soon as turning on \(S_{2}\), leading to higher switching-on losses. Moreover, both currents of the flyback and Class-D converters flow through the shared active switch (\(S_{2}\)). In practice, there is unavoidable leakage inductance in the coupled inductor of the flyback converter. When \(S_{2}\) is turned off, the current of the leakage inductance cannot be transferred to the resonant circuit. It will flow to and charge the parasitic capacitor of \(S_{2}\). Since the parasitic capacitor is usually very small, it will not only be charged to high spike voltage (as shown) but also cause high switching losses. Some lossless and passive snubber circuits have been reported to reduce the spike voltage. The resonant current in the series resonant current is shown in Figure 6e. Since the loaded quality factor is high enough, the resonant current is approximately a sine wave with a zero DC component. Figure 6f,g show the voltages and currents of the four LED strings, respectively. The measure voltages are 22.86 V, 23.18 V, 22.53 V, and 23.08 V while the measured currents are 0.75 A, 0.77 A, 0.76 A, and 0.76 A, respectively. The maximum difference of each string current is about 0.02 A, which should be the value of the magnetizing current of \(T_{2}\). The measured circuit efficiency is 88.2%. As predicted, the LED currents are almost the same; this proves the current balancing capability.