## 2 Structural Design and Analysis  
To decrease the structural dimensions and ensure the polarization-insensitive characteristic, a subwavelength cross-meandering resonator with a quadruple rotational symmetry was designed, as shown in Figure 1a. The resonator comprises a metallic layer and a dielectric substrate. The material of the metallic layer is copper because of copper's high conductivity (\(\sigma\) = 5.8 \(\times\) 10\({}^{7}\) S/m) [21; 22]. The dielectric substrate is set to F4B265, with a relative permittivity of 2.65 [23]. The thicknesses of the metallic layer and dielectric substrate are both 1 mm. Figure 1b displays a top view of the cross-meandering resonator, which is formed by rotating the meander gap structure by 90\({}^{\circ}\), 180\({}^{\circ}\), and 270\({}^{\circ}\), successively (Figure S1 of the Supplementary Materials). The resonator offers a long induction current path within a limited size because its design is a good combination of folded and slotted structures [24; 25], which can greatly reduce the resonator's dimensions. Here, the number of corners, which refers to the number of bends in the resonator branches (green line marks in Figure 1b), is \(N\) = 6. The size (\(w\) = 29 mm) of the resonant element is only 1/11 of the free-space wavelength, which is much smaller than that of previous designs [9; 11; 13; 19; 26; 27; 28]. To investigate the electromagnetic properties of the cross-meandering resonator, full-wave numerical simulations were carried out based on the finite element method. The structure was excited by a \(y\)-polarized electric point dipole 30 mm above the resonator. A schematic diagram of the simulation calculation is displayed in Figure S2a of the Supplementary Materials. The scattering boundary conditions and perfectly matched layers are applied on the sides of the structure. When the geometric parameters of the resonator are \(a\) = \(b\) = 1 mm, \(N\) = 6, and \(l\) = 15 mm, the electric field distributions (\(|E|\)) at the electromagnetic frequency of \(f=900\) MHz and \(f=1000\) MHz are as shown in Figure 1c. It can be seen that the field energy is mainly localized in the meander gap of the resonator at \(f=900\) MHz. The power ratio of the cross-meandering resonator is calculated by the formula \(G=10\times\log_{10}\left(P_{1}/P_{0}\right)\), where \(P_{1}\) and \(P_{0}\) represent the radiant energy flow with and without the resonator, respectively. The calculated power ratios at different frequencies are displayed in Figure 1d. A resonance mode at around \(f=900\) MHz is observed, and the 3 dB bandwidth is about 10 MHz. Because of the significant near-field interaction (about 180 times that at \(f=1\) GHz) [29; 30], the power ratio at the resonance frequency reaches 12 dB. When the vertical distance between the electric dipole and the resonator changes, the calculated power ratios at different frequencies are as shown in Figure S2b of the Supplementary Materials. It can be seen that the power ratio at the resonance frequency reduces with the increase of the vertical distance. The equivalent circuit model (ECM) is used to analyze the resonant properties of the cross-meandering resonator in theory. The transmission spectrum of the resonator array is shown in Figure S3 of the Supplementary Materials. There is a resonant peak at \(f=900\) MHz in the transmission spectrum, which also verifies that the resonator can effectively transmit a 900 MHz wireless communication signal. Next, the properties of the cross-meandering resonator with changes to its geometric parameters are investigated in detail. The calculated power ratios as functions of the length (\(l\)) and width (\(a\)) of meandering gaps, the width of the meandering strips (\(b\)), and the number of corners (\(N\)) are displayed in Figure 2a-d. According to the transmission-line theory [31; 32; 33; 34], the resonance frequency is inversely correlated with the equivalent inductance and capacitance of the cross-meandering resonator [35; 36].