## 2 Circuit Design and Performance  
The block diagram of the proposed microwave down converter is reported in Figure 1. The entire system was simulated within the Cadence AWR Microwave Office design environment. The 2-18 GHz weak received signal is amplified by means of the CMD240C4 low-noise amplifier (LNA) by Qorvo, Inc. Thereafter, the amplified signal is applied to the MM1-1467H mixer by Marki Microwave, Inc., with the aim of performing the signal up-conversion to the 1 GHz bandwidth centered on 29 GHz. This task is accomplished by exploiting the mixer sum output, avoiding too high LO signals and, thus, the use of highly expensive components or multiple multiplication stages, which would dramatically increase the spurious number. The LO generation is addressed by injecting the 13.5-5.5 GHz signal generated by a SW0220 microwave synthesizer by Kratos Defense and Security Solutions, Inc. into a frequency doubler stage MMD 1030 HS by Marki Microwave, Inc. The obtained 27-11 GHz signal multiplied by the 2 18 GHz input allows us to obtain a 29 GHz sum frequency with minimum spurious generation. Thereafter, the up-converted signal is amplified by the AMM 6702 amplifier by Marki Microwave, Inc. and filtered by means  
Figure 1: Block diagram of the dual–conversion microwave down converter. of the PB1182 WB bandpass filter by A1 Microwave Ltd. within the bandwidth of about 1 GHz. The power spectrum of the signal after the filtering stage is reported in Figure 2, showing how the in-band spurious signals are severely reduced by the effective filtering. The test simulations were performed by considering a sample input frequency of 6 GHz, hence generating a 23 GHz LO signal. It is worth noting that A1 Microwave Ltd. does not provide the scattering (S-) parameters for the PB1182 WB filter; thus, its behavior was synthesized by designing an equivalent lumped-element network. In detail, the main filter parameters were reproduced by implementing a Chebyshev topology by means of the well-known insertion loss method, which allows for a high degree of control over the in-band and stopband amplitude and phase characteristics [24]. A nine-elements network was chosen in order to obtain the desired matching with the real filter performance. In Figure 3a, the filter schematic is shown, while the insertion loss and return loss of the filter are reported in Figure 3b. The excellent matching between the real and reproduced curves might be verified by observing the data reported in the datasheet of the component [25]. The values of the filter elements shown in the schematic are reported in Table 1.  
Figure 3: (**a**) Schematic and (**b**) insertion loss (blue solid line) and return loss (red dotted line) of the simulated PB1182 WB filter. Figure 2: Power spectrum of the up-converted signal after the PB1182 WB filter. Thereafter, the MM1-1467H mixer is employed in the down-conversion stage. To this purpose, the APV-28-10 phase-locked oscillator (PLO) by Atlantic Microwave Ltd. (Braintree, UK) generates the 28 GHz fixed frequency for the second LO input. Such a LO frequency is required to obtain the mixer difference frequency output centered at 1 GHz, which is finally filtered by the VLF-1500+ filter by MiniCircuits. The intermediate frequency (IF) signal might be easily processed by employing currently available analog-to-digital converters. The IF signal spectrum at the output of the VLF-1500+ is reported in Figure 4, where the high spurious suppression is visible. The down-converting chain components were carefully selected by paying great attention at the recommended LO power levels and at matching the dynamic range of adjacent components. The power budget of the down converter with line marks identifying the inter-stage output 1 dB compression points, OP1 dB, is reported in Figure 5.