## 1 Introduction
Indeed, under the assumption that the rise time equals the fall time, i.e., \(t_{\text{r}}=t_{\text{f}}\) the envelope of a trapezoidal waveform is characterized by two break frequencies, at \(f_{\text{b1}}=1/(\pi\tau)\) and \(f_{\text{b2}}=1/(\pi t_{r})\)[6]. Since typically \(t_{\text{r}}<<\tau\), it is \(f_{\text{b1}}<<f_{\text{b2}}\), meaning that \(f_{\text{b2}}\) determines the high-frequency spectral content of the CM source voltage, and of the CE, eventually. Actually, the switching voltage (\(v_{\text{sw}}\)) can be affected by oscillations taking place during the switches turn-on and turn-off. More precisely, such oscillations are triggered by voltage or current steps exciting resonant circuits, which comprise the capacitive and the inductive parasitics of the power switches themselves, of the package interconnections and of the printed circuit board (PCB) layout. As a consequence, peaks at the resonant frequencies can appear in the \(v_{\text{sw}}\) frequency spectrum, as well as in the \(i_{\text{CM}}\) one, eventually. The superimposed oscillations also affect radiated emission, as reported in [7, 8], thus they must be definitely avoided. Standard solutions to reduce the amplitude of such oscillations include the use of resistor-capacitor (RC) snubbers [9], ferrite beads or to slow down transients by means  
Figure 1: Conducted emission setup for a single leg. The differential- mode (DM) and common-mode (CM) currents are highlighted, as well as their major sources, i.e., the switching current (\(i_{\text{sw}}\)) and the switching voltage (\(v_{\text{sw}}\)), respectively. of a higher output resistance of the driver. Although the aforementioned techniques are effective in avoiding such unwanted oscillations and in reducing the delivered EMI by decreasing the \(\mathrm{d}\mathrm{V}/\mathrm{d}\mathrm{t}\)[10], on the other side, the switching losses of the power switches increase significantly, resulting in a lower efficiency of the power circuit [11, 12]. For such a reason, new solutions to find a better trade-off between EMI reduction and efficiency are currently being investigated. Amongst them, the use of active gate drivers (AGD) has been proven to be effective in controlling the switching trajectories of power transistors, resulting in a better compromise between overshoot reduction and switching power losses [13, 14, 15]. The key idea of AGDs is to modify the strength of the driver during the turn-on and the turn-off of power transistors, more precisely, it is required to slow down the transistor in the middle of the commutation, while keeping fast the remainder part of the transition [16]. In such a way, the transistor itself is exploited as a dissipative element, meaning that instead of driving it from a very high resistance, i.e., the off state, to a very low resistance, i.e., the on state, it is driven to modulate its output resistance during the commutation to avoid the oscillations from taking place. However, AGDs suffer from some major cons, as they require a complex hardware, they are not commercially available, and their tuning may be challenging to achieve if no feedback is exploited [17]. The switching waveforms obtained using an AGD can be closely reproduced using a simpler and cheaper passive component available on the market. The key idea of the work presented in this paper is to modulate the driver strength of a high-side power transistor without using an AGD, but exploiting the source inductance. Indeed, during the transistor turn-on, such an inductance couples the power loop with the driver loop, so that the control voltage of the transistor is reduced during the transient, as in the case of an AGD. As a consequence, the switching waveforms are differently shaped with respect to a conventional gate driver, and the oscillations resulting from the excited parasitic resonant circuit are reduced.