esa-sceva/satcom-chunk-collection huggingface dataset:
## 2 Materials and Methods
### Feature Extraction
The overlap associated with the periodogram calculated with the Welch method helps to significantly improve the correlation of the data, improving its statistical properties and therefore improving the reliability of the estimate [37]. For discrete-time systems, the Welch periodogram returns an estimate of the power spectral densities. This method segments the time signal into periods of a certain block size. For each period, the spectrum is calculated through an average estimate of the final spectral density. The block size affects the duration of the individual segments and the spectral resolution of the final estimate of the power density. The PSD extracted from the recorded signals was subsequently used for the estimation of the magnitude-squared coherence. These values were then used as input for the ANN-based model. To evaluate the differences between two signals, it is possible to use the concept of coherence which represents an ideal property of waves: It allows a stationary interference that is constant in time and space. Two wave sources are perfectly coherent if their frequency and waveform are identical and their phase difference is constant. More generally, coherence describes all the properties of the correlation between physical quantities of multiple waves. So, to evaluate the effects of the room on the sound field we can use the coherence between the recorded signals (using the Zoom H6 Handy Recorder (ZoomCorporation, Tokyo, Japan) equipped with a X-Y microphone) and the anechoic signal. Coherence is a function of the power spectral densities of the two signals, and of the cross power spectral density [38,39]. Cross-power spectral density is a statistical property of a pair of random sequences that indicates the likely distribution of power over different frequencies at any given time. The cross-power spectral density of a pair of X and Y signals is the Fourier transform of the cross correlation between the signals. Given two signals \(a\) and \(b\), the coherence function is defined by Equation (7):  
\[Coh_{a,b}(f)=\ \frac{\left|PSD_{a,b}(f)\right|^{2}}{PSD_{a,a}(f)\times\ PSD_{b,b}(f)} \tag{7}\]  
Here,  
* \(PSD_{a,b}(f)\) is the cross power spectral density for \(a\) and \(b\) signals
* \(PSD_{a,a}(f)\) is the power spectral density of \(a\) signal (coherent signal)
* \(PSD_{b,b}(f)\) is the power spectral density of \(b\) signal (no-coherent signal = signal modified by the reverberation of the room)
# Detection and Localization of Overlapped Fruits Application in an Apple Harvesting Robot
## 4 Overlapped Circles Localization
### Centers Detection  
The outline of the apple can be regarded as a quasi-circular, so the circle fitting method can be used to recognize and locate the apple target. For circle fitting, it needs to find the two most basic elements: the center and radius of the circle. The minimum distances from the point in the apple  
Figure 5: Morphological processes: (**a**) image smoothing; (**b**) binary process; (**c**) corrosion; (**d**) dilation; (**e**) holes filling; (**f**) contour extraction.  
contour to the edge of contour, the maximum value corresponding to this point in these minimum distances is the center of the circle, that is to find the local maximum value of the minimum distance function to find the center of the circle. The key is to get the above function. If the distance is calculated directly, it will generate a large amount of data, thus affecting the real-time capability of the system. Therefore, the fast generating distance function method is used to solve this problem. The program will perform four scanning processes for the pixels within the outline, and then the minimum distance of all inner pixels will be found accurately and quickly.  
Four scanning directions should be defined first:  
(x+, y+): from left to right, from top to bottom; (x-, y+): right \(\rightarrow\) left, top \(\rightarrow\) bottom; (x+, y-): left \(\rightarrow\) right, bottom \(\rightarrow\) top; (x-, y-): right \(\rightarrow\) left, bottom \(\rightarrow\) top.  
The whole process is as follows:  
Initialization: The distance value between the outer and edge points is defined as \"0\" and marked as \"Visited\". While the distance value of the internal point is defined as '\(+\infty\)' and marked as \"Unvisited\".  
Scanning: Scan the outline of the apple in four defined directions. For each time, the distance value of each pixel E will be updated. The updating method is as follows:  
a. (x+, y+) direction: For each internal pixel E(i, j), compare the distance values of its left pixel E (i-1, j), and top pixel E (i, j-1), and then add 1 to the smaller value (definition For'min' ). If min +1 is less than the distance value of E (i, j), update min + 1 to this value. Otherwise, the value remains unchanged.  
Otherwise, the value remains unchanged. Similarly, the three directions of (x-, y+), (x+, y-), (x-, y-) are scanned separately.  
After the scanning process, each internal point is compared with its four neighboring pixels and the distance value reaches the minimum value. Then, the distance value of each pixel is combined into a matrix. The 3D map of this matrix is shown in Figure 6.  
It is obvious that there are two local maxima pixels, which correspond to the two circle centers in the overlapped apple images. Take these two coordinates, and they are the center of the circle of the overlapped apples' contour. The numerical value in the corresponding distance function matrix of coordinates is the requested radius, so the apple positioning has been achieved.
## 5 Applications of PCMs in Buildings
### PCM Panels and Wallboards
#### 5.1.2 Floor Applications  
Building floors are another area where PCM technology can be applied for energy saving by thermal regulation. The applications that have been investigated for floor solutions are very varied. The addition of PCMs in the construction can be by adding a single layer of PCM [100; 101] or by adding a multilayer of different types of PCMs [102; 103]. Additionally, another approach is the development of floor panels with capillary network for PCM circulation [104], or the use of PCMs along with an electrical powered system for diminishing the demanded energy consumption [105]. Finally, PCMs can also be integrated in the raw materials used for construction, such as concrete [91]. Wanchun Sun et al. [84] investigated the thermal performance of a double-layer PCM radiant floor system containing two types of inorganic composite PCMs. The composites PCMs used were Na\({}_{2}\)HPO\({}_{4}\)\(\bullet\)12H\({}_{2}\)O@EG and CaCl\({}_{2}\)\(\bullet\)6H\({}_{2}\)O@EG. During winter, the thermal comfort lasted 2.2 times more than the reference room, which contained pebbles in the floor. During the summer, the thermal comfort lasted 1.7 times more. In both cases, the system can achieve shifting of the peak load. Shilei Lu et al. [86] studied the manufacturing of a PCM floor heating system consisting of a double pipe with three different modules. Regardless of the module, indoor thermal comfort was achieved. The average indoor temperature fluctuation was 1.8-3.0 \({}^{\circ}\)C.
## 2 Materials and Methods
### Extraction and Analysis of Volatile Organic Compounds by SPME-GC/MS  
The extraction of volatile organic compounds (VOCs) was performed using Headspace-SPME technique according to the procedure described by [PERSON] et al. [PERSON] et al. (2018). In brief, 10 mL of sample were transferred into a 20 mL glass vial, and 50 \(\upmu\)L of 2-octanol (purity 99%, Sigma-Aldrich, St. Louis, MO, USA) were added as an internal standard (178 mg/L in ethanol solution). The vial was conditioned at 50 \({}^{\circ}\)C for 5 min to allow the sample to equilibrate and to favour the partitioning of VOCs between the matrix and the headspace. VOCs were adsorbed by inserting a 50/30 \(\upmu\)m divinylbenzene/carboxen/polydimethylsiloxane (DVB/CAR/PDMS) 2 cm fibre (Supelco, Bellefonte, Pennsylvania, USA) into the headspace of the vial, exposing the polymer for 15 min at the same temperature.  
VOCs were desorbed directly in the injector port of GC kept at a temperature of 250 \({}^{\circ}\)C in split mode with a 4:1 split ratio, for 10 min. Volatile compound analysis was performed on an Agilent 7890A GC System gas chromatograph coupled to an Agilent 5975C VL MSD with Triple-Axis-Detector mass spectrometer (Agilent Technologies, Inc., Palo Alto, CA, USA). GC was equipped with a Zebron ZB-WAX capillary column (60 m \(\times\) 0.25 mm i.d. \(\times\) 0.25 \(\upmu\)m film thickness 100% polyethylene glycol; Phenomenex, Torrance, CA, USA). The carrier gas was helium with a flow of 1.3 mL/min. The temperature program was as follows: 40 \({}^{\circ}\)C for 5 min, then ramped at 2 \({}^{\circ}\)C/min to 220 \({}^{\circ}\)C, and held at the maximum temperature for 20 min [PERSON] et al. (2018). Mass spectra were recorded at 70 eV. The source temperature was 230 \({}^{\circ}\)C, the quadrupole temperature was 150 \({}^{\circ}\)C and the interface temperature was 250 \({}^{\circ}\)C.  
The identification of VOCs was performed by comparing retention times and mass spectra obtained by analysing pure reference standard compounds (Sigma-Aldrich, St. Louis, MO, USA) in the same conditions. The identification was confirmed by comparing mass spectra with those of the National Institute of Standards and Technology (NIST) database. The fibre was conditioned at 270 \({}^{\circ}\)C for 1.5 h before the analysis. A blank test was performed before each analysis. The quantitative data of the volatile compounds were obtained by normalising the peak areas of each compound with respect to the peak area of the internal standard. Peak area data were processed by MSD ChemStation 5975 TAD Data Analysis software v.E.02.00.493 (Agilent Technologies, Palo Alto, CA, USA). Analyses were conducted in triplicate.
# Efficient High-Dimensional Quantum Key Distribution with Hybrid Encoding
## 1 Introduction
We evaluate the secret key rate of our scheme with respect to the experimental parameters and identify the regime where our scheme is more secure than the original DDI-QKD. In addition, we also compare the security of our scheme with that of high-dimensional MDI-QKD (for the case of \(d=3\)). We note that our protocol is more secure than the original BB84 protocol against a side channel attack (but less secure than MDI-QKD). For example, it can detect the basic detector blinding attack [12] from double clicks of detectors [28; 29]. On the other hand, the attained key rate with our protocol is comparable with BB84 protocol, while MDI-QKD has a half of signal sifting rate of BB84 protocol due to the 50% limit of the success probability of the BSM. Although DDI-QKD is not as secure as MDI-QKD and requires trusted elements in BSM setup, the main idea of employing two different DoFs motivated by DDI-QKD still merits consideration for practical usage in some secure communications e.g. quantum secret sharing [51]. As we demonstrate in this article, it is possible to improve the security as well as the efficiency over the original DDI-QKD in high-dimensional approach. In addition, we here propose a feasible high-dimensional QKD scheme with OAM of a single photon, while a high-dimensional MDI-QKD may be hard to realize due to the difficulty in implementing high-dimensional BSM on two photons with linear optical elements [19]. This article is organized as follows. A schematic description of the \(d\)-dimensional QKD (\(d\)-QKD) with hybrid encoding is presented in Section 2, and its practical implementation is in Section 3. In Section 4, we analyze the secure key rate of our protocol. Finally, conclusion on the efficiencies is drawn in Section 5.
## 2 Preliminaries
### Bilinear Map  
We briefly review the necessary facts about bilinear maps and bilinear map groups. Let \(G\) and \(G_{T}\) be two cyclic multiplicative groups of the same prime order \(q>2^{\lambda}\), where \(\lambda\) is the security parameter, and let \(g\) be the generator of the group \(G\). We define that \(e:G\times G\to G_{T}\) is a bilinear or pairing map if it satisfies the following conditions:  
* Bilinear: For all \(a,b\in Z_{q}\), \(g\in G\), \(e(g^{a},g^{b})=e(g,g)^{ab}\);
* Non-degenerate: \(e(g,g)\
eq 1_{G_{T}}\), i.e., if \(G=<g>\), then \(G_{T}=<e(g,g)>\);
* Computable: The map \(e\) is efficiently computable. Generally, the map \(e\) is obtained from Tate or Weil pairings.
# Experimental Study of Radiated Emission Due to Secondary Common-Mode Current in Buck Converters
## 3 Generation Mechanism of the SCM Noise in the Experimental System
For the experimental system in this article, it is necessary to connect an artificial impedance network between the ground line terminal E and the ground plane to achieve the balance of power supply side impedances. However, in practice, the power supply side impedance is not constant over a wide bandwidth. Generally, it is difficult to accurately simulate the frequency characteristics of a power supply side impedance over wide-band frequencies. Moreover, from the viewpoint of preventing electric shock, it is undesirable to connect an impedance balancing component to the ground line of power converters. On the other hand, the PCM current can be suppressed using CM inductors (CMIs) to increase the impedance of the PCM propagation path [20; 34]. The following section demonstrates the above-mentioned radiated noise generation mechanism by connecting CMIs to the experimental system and evaluating the impact of CMIs on each CM current and radiated noise based on the experimental results.
# Data Security-Based Routing in MANETs Using Key Management Mechanism
## 4 Simulation Environment and Performance Metrics
### Packet-Delivery Ratio  
The packet-delivery ratio is the ratio at which the data packets are successfully delivered to the destination. It can be computed as:  
\[\text{PD}=\frac{\sum_{\forall i\in D}TPR_{i}}{\sum_{\forall i\in D}\times TPS _{k}}\times 100 \tag{13}\]  
where \(TPR_{i}\) represents the total number of packets received by the \(DBR\) destination, \(i\) and \(TPS_{k}\) represent the total packets sent by the Constant Bit Rate (CBR) source \(k\), and S denotes a collection of CBR sources. \(D\) represents a collection of CBR destinations.
# Discontinuous Yielding of Fe E420 under High Strain Rate Loading
## 1 Introduction
[15] demonstrated that both pre-straining and bake hardening treatments can induce a DY behaviour in two CMni transformation-induced plasticity (TRIP) steels. They concluded that it is due to the treatments-induced micro structural changes, such as an increase in the dislocation density and the formation of fine precipitation in bainite and martensite. The strain-rate dependency of a dislocation pinning dependent DY was observed by [PERSON] and [PERSON] [16] in centre-annealed wires of three alloys Ni3 Mn, Ni3 Fe and Ni3 Fe-5%. The DY phenomenon in Ti-40 at.% Al alloy, was investigated by [PERSON] et al. [17], at strain rates in the range of \(1\div 0.001\) s\({}^{-1}\). The ratio between the upper and lower yields was found to increase with the strain-rate. Further, a strong correlation between the microstructural evolution and the DY in the dependence of the strain-rate was found. In contrast, a negligible upper yielding was observed for the thermally treated Al-2024 alloys. It was associated with a flow stress decreasing with the strain-rate in fully annealed material, while an opposite trend was found in fully aged material [18]. [PERSON] et al. [19] studied the DY in ultra-fine-grained austenitic stainless steels under temperature. They found no systematic dependence on the Luders strain on the strain-rate and a drastic increase of the plateau stress with the temperature decrease in 316 steel. Since its dependence on material micro-structure, dislocation and interstitial atom density, as well as thermal treatment, temperature and strain-rate, the DY characterization is complex and still not a completely unravelled phenomenon. For this reason, the main aim of the present work is to investigate the DY of the quenched Fe E420. It is a low carbon steel widely adopted in automotive and transportation fields, both for structural and non-structural components. Since these components may be subjected to static and dynamic loads over their life cycle, a comprehensive characterization of the material response under the described conditions produces results of great interests. For these reason, in the present work, the DY behaviour of both thermally treated and as-is Fe E420 steel was investigated under three different strain rates equal to \((2\times 10^{-4},500,\) and \(1000)\) s\({}^{-1}\).
## Appendix J Description of Mechanisms for Telecom  
### Delegated Proof-of-Trust (DPoT) [574]  
Lwin et al. [574] propose a consensus mechanism for mobile ad hoc networks (MANETs). The proposed scheme is a trust management system employing a combination of policy-based and reputation-based trust management. The reputation component is anchored in trust values, a metric that determines whether a node is trustworthy based on the node's past behaviour. The policy component is implemented through a series of rules that dictate how and when trust values are updated. To validate blocks in the proposed scheme, a DPoT algorithm is used: following this, nodes with high trust values are elected as validators. While the reputation system can prevent small-scale Sybil attacks, well-orchestrated ones are unlikely preventable due to the possibility of a large number of Sybil identities colluding to artificially inflate their score.
# Analysis and Design of Integrated Blocks for a 6.25 GHz Spacefibre PLL
## 2 System-Level Design of PLL and Passive Filter Sizing  
The target architecture of this work is a CP-PLL [18]. As shown in Figure 1, the blocks that compose the proposed PLL are a PFD, a CP circuit, a passive Loop Filter, a VCO, whose output corresponds to one of the PLL outputs, and an FD with an integer divide ratio, which generates the other three frequencies required by the application. The input signals of the PFD are the reference signal (REF in Figure 1) and the FD output, while the outputs related to the input phase/frequency difference are named UP and DOWN in the figure. These signals are converted by the CP in a charging or discharging current needed to control the VCO through the Loop Filter. The PLL proposed in this work is designed starting from an LC-tank VCO macrocell, which is integrated by the University of Pisa in 65 nm CMOS technology [16] and has the following characteristics: a frequency tuning range between 5.9 and 7.5 GHz, obtained through integrated varactors; the control voltage leads to a gain in the center of the frequency range of 2 GHz/V; the Phase Noise is below \(-100\) dBc/Hz at 1 MHz from the carrier.  
Starting from the system-level analysis and design, the Loop Filter's components, the Charge Pump's current (I\({}_{\text{cp}}\)), and the Frequency Divider ratio (N) have been chosen. Thanks to a divider ratio of 40 and with 156.25 MHz as the reference frequency, the output at 6.25 GHz is obtained. According to the Spacefibre standard, the PLL has also to generate the frequencies 3.125 GHz and 1.5625 GHz, which are obtained using an integer FD with the following ratios: one-half, one-fourth, and 1/40 th.  
Moving onto the Loop Filter, as shown in Figure 2, it is composed of a capacitor C1, a resistor R1, which is needed to stabilize the loop, and a capacitor C2, whose aim is to reduce the spurious tones at multiples of the reference frequency and whose value, according to [19], should be no more than C1/5. Thanks to the modeling and simulation activity in the ADS environment in Section 2.1, a bandwidth of 6 MHz has been targeted as a tradeoff between the noise behavior and Loop Filter integrability. A completely integrated filter allows the avoidance of all the problems deriving from the parasitic effects of external components. A higher loop bandwidth, obtained with small passive devices, provides a better reduction of the VCO and Loop Filter contribution to Phase Noise. Instead, a lower bandwidth leads to lower CP and reference noise contributions, but it can be achieved only with large capacitors. A lower CP noise contribution can also be obtained thanks to a higher \(\mathrm{I_{cp}}\), at the cost of larger capacitors for a given bandwidth. Given all these reasons, an \(\mathrm{I_{cp}}\) of 40 \(\mathrm{\SIUnitSymbolMicro A}\) has been chosen, and consequently, the values of C1, C2, and R1 have been derived: 8 pF, 1 pF, and 12 k\(\Omega\), respectively.
# A Compact, Low-Profile, Broadband Quasi-Isotropic Antenna for Non-Line-of-Sight Communications
## 2 Broadband Quasi-Isotropic Antenna Design
### Development of a Fork-Shaped Crossed Dipole Antenna  
The development stages of the proposed antenna are shown in Figure 4. Note that the radius of the substrate and the radius and width of the vacant ring remains the same for all development stages. An ideal port, and not a modeled coaxial cable, is utilized in this section and in the parameter studies. A conventional crossed dipole, as shown in Figure 4a, is popular for its relatively easier form factor to tune the polarization and radiation pattern synthesis of unidirectional or quasi-isotropic patterns. However, the desired pattern could deteriorate as a result of bandwidth enhancement due to the non-uniformity of the current distribution on the radiating surface.  
For example, when the shape of the dipole arms is tapered to enhance the working bandwidth, as shown in Figure 4b, the current is mainly concentrated along the periphery (or in the \(\phi\)-direction) of the tapered arms, and it negatively affects the radiation performance. Note that the central portion of the tapered arms in Figure 4b, where the surface current density is relatively weak, can be removed for current uniformness, with a negligible effect on the impedance bandwidth. This results in a fork-shaped proposed crossed dipole antenna, as shown in Figure 4c. As illustrated in Figure 4c, the direction of the current distribution is more aligned along the dipole arms than that in the case of Antenna II, where the current distribution in the \(\phi\) direction is stronger, as shown in Figure 4b.  
The three arms In the proposed fork-shaped design can create resonances at three different frequencies, which is beneficial in enhancing the impedance bandwidth. Overall, the proposed antenna in Figure 4c shows a broad impedance bandwidth from approximately 1.8 to 4.3 GHz, with a fractional bandwidth of 81.9% with a \(-\)10 dB criterion, as shown in Figure 5a. As shown in Figure 5b, the gain deviation level and its performance bandwidth are improved when compared with Antennas I and II, clearly indicating the better performance of the crossed dipole design. However, it is also worth noting that the  
Figure 4: Development of the proposed antenna design and its current distribution at 2.4 GHz. (**a**) Original crossed dipole antenna (Antenna I). (**b**) Tapered-shaped crossed dipole antenna (Antenna II). (**c**) Proposed fork-shaped crossed dipole antenna.  
Figure 3: Configuration of the proposed fork-shaped crossed dipole antenna. (**a**) Top view. (**b**) Side view.  
total gain is lowered compared to that of Antenna II since part of the radiating patch is notched out. This can be regarded as a tradeoff between the maximum gain and the gain flatness across the frequencies.