>
V
I
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20 June 1980
] Quarterly Reports
1 &
m
Prepared by
]
E,
C #
May
^
G.
S. Hubbard
(consultant)
333 Ravenswood Ave. • Menlo Park, California 94025
(415) 326-6200 • Cable: SRI INTL MPK
•
TWX: 910-373-1246
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ABSTRACT
We have constructed
a computer-based random number generator and
analyzer system.
Two sources
of
true random input were used
(a P-decay
source and an electronic noise
diode),
and special attention was
devoted
to insuring their
independence from environmental influences.
Extensive
testing has
shown that
the system performs according to expectations,
and
a
data sample
of approximately 10® bits
from each
source meets all appro-
priate
criteria
for accepted definitions
of randomness.
iii
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CONTENTS
ABSTRACT.
iii
LIST OF
ILLUSTRATIONS
LIST OF TABLES.
.
.
.
EXECUTIVE SUMMARY
.
I
SYSTEM DESCRIPTION
A.
Sources
1.
Noise Diode
2.
3-Decay
3.
Hardware Pseudorandom Shift Register
.
.
.
B.
Analysis and Control
1.
Digital Equipment Corporation
(DEC) LSI-11
Microcomputer
2,
Software
C.
Display
1.
Grinnell Graphics Display System
2.
Summagraphics
Tablet
II
SUBSYSTEM TESTING
A.
Noise Diode RNG
1.
Device Tests
,
,
.
#
2.
System Isolation and Monitoring.
B.
3-Decay RNG
1.
Device
Tests
2,
System
Isolation and Monitoring.
C.
Computer Pseudorandom Generator
.
.
.
III
SYSTEM ANALYSIS.
iv
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IV
CONCLUSIONS
Appendices
A
TECHNICAL CONSTRUCTION DETAILS OF RANDOM SOURCES. ...
B
TECHNICAL DETAILS OF SPECIAL INTERFACE BOARD
REFERENCES
v
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ILLUSTRATIONS
Block Diagram of Computer-Based RNG System
Pulse-Processing Sequence
A Texas
Instruments
MD—20 Noise Diode
is
Shown with
Its
Associated Electronics
#
0-Decay from a
147Pm Source
is Detected by a Surface-Barrier
Radiation Detector
to Provide
a Random Signal
Slot Diagram—Front
View
Example
of
Sequential
Sampling
Plot,
Pulse Height Distribution
of Filtered Diode Noise from -40
to +40° C
Pulse Height Analysis
of Diode D-l
at
Two Leakage Current
Values
. ......
Representation
of
Single-Sweep Pulse Train from Noise Diode
at
a
Current
of
150 pA
Log
of
the Pulse
Rate
as Measured by Frequency Analysis
Plotted
as
a Function of Band Gap
Battery Voltage and Current
as
a Function
of
Time
Diode Temperature versus
Time
Detector
Leakage Current
as
a Function of Temperature.
.
.
.
vi
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TABLES
Pseudorandom Sequence from 8-Bit
Shift Register.
•
•
Subcomponent Manufacturers
Address Assignments
Results
of Statistical Tests
of
the Random Number
Generators
*
Sequential Analysis
of Random Sources
in Sets
of
100
Trials
B-l
Definition of Command-Byte Bits. ...
vii
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EXECUTIVE SUMMARY
General
The purpose
of
this program is
to develop
a reliable
computer-based
random number generator
(RNG)
system with graphics
display capabilities.
The system is designed to reduce to acceptable
levels,
or to eliminate
entirely, external interference with the random source elements.
In order to achieve
the objective
of this
program, we have developed
a
computer-based random number generator.
Special efforts
have been made
in two specific areas:
First,
extensive testing
of
the
true random sources
was
carried out
to study their response to environmental
factors.
Second,
a variety of
statistical tests
have been applied to the complete
system
in order
to ensure that
the output
is truly random under experimental
conditions
,
We have assumed that the
computer and peripheral electronic equipment
will operate
in conformance with manufacturer's
specifications
and there-
fore will
not
require additional testing.
System Description
Figure
shows
the overall system design.
Random Sources
The random source elements
consist
of
a
commercially available noise
diode,
a radioactive source with an appropriate radiation detector,
and
a pseudorandom shift register.
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V
RANDOM
SOURCES
ANALYSIS + CONTROL
DISPLAY
FIGURE
BLOCK DIAGRAM OF COMPUTER-BASED RNG SYSTEM
A Texas
Instruments MD-20 planar silicon noise diode was
chosen for
its
large noise output
(-500 UV//Hz) and
its well-described functional
2 *
characteristics.
’
147Promethium was
selected as
a radioactive source because
it
is
nearly a 100% 3-emitter with
no competing decay modes.
Detection of the
electron continuum
is accomplished using
a well-understood and reliable
ORTEC silicon surface-barrier detector.
To act
as
a control
noise source
a standard pseudorandom shift
register was
constructed.
The binary output
of such a
device has the
property that although the sequence meets
a number of criteria for
randomness,
the sequence
is deterministic,
once the starting seed for the
register
is
given.
Figure
2 shows
the process
by which
a random number
is generated from
the noise diode source.
Random-amplitude
1-MHz sawtooth voltage pulses
from the diode
(Caption
1,
Figure 2) are filtered by a bandpass filter
(2).
*
References
are
listed at
the end of
this report.
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FIGURE
PULSE
PROCESSING
SEQUENCE
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At
each positive-going zero crossing
of the filtered signal
a TTL pulse
is generated, giving
a random digital signal
(3).
Finally,
a divide-by-
two circuit
changes
state at
the rising edge of each TTL pulse,
yielding
a binary bit stream
(4) with probability of being in the
logical
1 state
of
1/2.
This
bit
stream
is sampled and shifted into an 8-bit shift
register at
a
1-kHz rate,
so that
a random 8-bit number may be selected
at
intervals
greater than
ms
.
A completely analogous
process occurs
with the 6-decay source.
The major distinctions
are
that electrons
of
random energy arrive
at
a detector where they are converted into electrical
signals
of random voltage.
A low-level discriminator generates
a TTL pulse
whenever the voltage
rises above
a
certain point.
From this
point,
the
signal processing
is
the
same
as
described above.
Analysis
and Control
The analysis
and
control portion
of
the system consists
of an LSI-11
microcomputer
.
The LSI- 11
is
programmed to sample one
of the noise sources
at
a specified rate
to obtain its random bits.
A sequence
of
such samples
is tested by the LSI-11
for an excess
or deficiency of
l's
on a continuous
basis, using
a
sequential analysis statistical
technique. 3
’ 4
Sequential
analysis
is extremely efficient
for determining whether or not
the
output
of
the binary random generator
contains
a distribution of
0's and
l’s
as
expected.
The principal advantage
of the sequential sampling technique
as
compared with other methods
is
that,
on the average,
fewer bits per
final decision are required
(roughly 50% fewer)
for an equivalent degree
of statistical reliability.
Display
The computer-driven graphics display system consists
of two independent
19-inch color video monitors,
a Grinnell display controller,
and a Summa-
graphics 20-by-20-inch graphics
tablet.
Using these components,
data from
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sequential sampling statistics,
pulse height analysis,
or any other output
may be displayed.
System Testing
Noise diodes
for use
in this system were extensively tested for
response
to
changes
in temperature
(-40 to +40°C),
leakage current
(40 pA
to 200 uA)
and other environmental factors
such as
a 6000-gauss
dc
241
147
magnetic field and
low-intensity radioactive sources
(
Am,
Co,
PmJ
.
We found that over the
range examined for each factor the spectral noise
density was
flat within
±1
dB for the bandpass
of the filter
(1 kHz to
200 kHz).
Furthermore,
the filtered noise followed a gaussian distribution
under all
conditions tested
as
long as
the
leakage current was 80 to 120 pA.
.
147
.
.
The random emission of electrons from the P-decay of
Pm is inde-
pendent
of known external influences.
The element that
is sensitive
the
surface barrier detector—was
tested for
changes
in leakage current as
a
function of
temperature.
At the maximum temperature tested
(~40°C)
it was
found that
the noise
contribution due to the increased leakage current
could be
completely eliminated with an appropriate
low-level discriminator.
We assume that the TTL logic circuitry and devices
composing the shift
register conform to the
low fail rate found in the manufacturer's specifi-
cations,
thus eliminating the
need for further testing.
We have also assumed that
the other system elements
(LSI-11, Grinnell
controller etc.) will continue to operate as specified by the vendor
so
that extensive environmental testing
of these components
is unnecessary.
System Isolation and
Interference Protection
To prevent spurious
signals
due
to known external influences from
being incorporated
into the random source output,
numerous precautions
have been taken.
Each random source
is encased in a sealed
0 . 125- 1 nch-thick
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soft
iron box with radio frequency shielding, providing protection against
mechanical,
magnetic,
or RF intrusion.
Power
is supplied by batteries
to
eliminate ac line transients and 60-Hz noise.
All
data output
to the
LSI-11
is
via optical
links
to ensure
complete electrical
isolation.
In
addition,
the temperature of the noise diode
is
continuously monitored.
circuits
have been included in both random sources
so that
the units will automatically shut
off and must
be manually reset under the
following circumstances:
•
The battery supply drops
below
a critical
point.
•
The electron detector
leakage
current rises
above
an acceptable
level.
•
The
diode
current
deviates from a
narrowly defined-
current window.
Results
A variety
of fixed-length statistical
tests have been applied to
500,000 sample
control
runs
of random numbers
generated by the system
described above.
No unexpected deviations from chance expectation were
observed in these
control
runs,
indicating that
the system performs
in
accordance with design.
The numerical results are tabulated in Section
III.
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I
SYSTEM DESCRIPTION
A
.
Sources
.
Noise Diode
Commercially available random noise diodes typically have an
output
of ^ 10 pV /[Hz
,
An exception
is
the Texas
Instruments MD—20 noise
diode, which produces
a noise voltage on the order of
500
pV//Hz.
This
large noise voltage
is achieved by
a. device design
in which internal field
emission initiates avalanche breakdown bursts resulting in
~1 MHz random
amplitude sawtooth pulses.
This
diode
is incorporated into
a random number
generator
as
shown in Figure
3.
Sawtooth pulses
from the
diode are fed into a
6-pole bandpass
Butterworth filter with rolloff points
at
kHz and 200 kHz,
respectively.
Filtered output
goes to
(1)
an optical transmitter where
the analog data
are sent
to the LSI-11,
and
(2)
a zero-crossing detector.
The zero-crossing
circuit generates
TTL logic pulses
,
creating
the digital Poisson distributed
output.
This binary string
is
also transmitted to the LSI-11 via an
optical
channel.
A platinum resistor
has
been bonded to the
noise diode
to provide temperature sensing.
The resistor’s temperature coefficient
is extremely well
characterized
(T
= 0.385 Q/°C)
and quite
stable.
Using
a voltage-to-f requency converter,
the temperature
data are transmitted to
the
LSI-11
via
a
third optical
link.
Power
is
supplied from rechargable
Ni
-
Cd batteries
.
In its
current
configuration the unit
can operate
for approximately 5.5 hours
before the supply voltage
drops below the
cutoff point.
Recharge time
is
10 to
15 hours.
Current sensing circuitry will
shut
off
in the
system
if
the diode operating point
falls below 95 pA or rises above
105
pA.
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FIGURE 4
(3-DECAY FROM A
SOURCE
IS DETECTED BY A SURFACE-BARRIER
RADIATION DETECTOR TO PROVIDE A RANDOM SIGNAL.
All
output
is
via
optical
signal
coupling.
coupled output
for analog pulse height analysis,
and
(2)
a TTL pulse
generator.
Each event greater than 25 keV creates
a
logic pulse.
The
output
is
a random binary string,
Poisson distributed
in time,
just
as
in the noise diode RNG
.
These binary pulses are also sent
to the computer
by an optical
channel.
Power
is
supplied
in the same way as
for
the
noise diode
RNG.
If
the detector
leakage
current
rises above
pA,
the system shuts
down.
3 .
Hardware Pseudorandom Shift Register
A third source
of random input
is that which
is
generated from
a pseudorandom shift register.
The
sequences
derived from such devices
are well understood,
5 and the randomness properties
can be specified
exactly.
For example,
if
a maximum- length binary sequence
is
chosen,
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there will be one
less excess number
of binary
l
f s than the expected
number, which is
1/2
the sequence
length.
Also,
the
length
of the runs
follows
a
decreasing geometric progression,
in that
the probability
of
finding
a run of
length n +
is
half
that
of finding a run of
length
n.
Lastly,
the autocorrelation function of the
complete sequence
is only
two-valued,
for the zero-lag and nonzero-lag
cases respectively.
To serve
as
a form of
control, we have chosen an 8-bit register
with the generating polynomial given by
X
+ X
+ X
+
Starting from the initial value
of 255,
255 pseudorandom numbers
are
generated
in the order shown in Table
1.
This pseudorandom generator resides on the general interface
board and
can be initialized and examined under software
control.
B
.
Analysis and Control
.
Digital Equipment
Corporation
(DEC) LSI-11 Microcomputer
The LSI-11 microcomputer
is
a standard item manufactured by DEC.
It
is
chiefly characterized
by its
internal
bus structure and ease
of
operation from both the hardware and software perspective.
Because
of
these
latter two points,
LSI-11
systems have gained
a wide acceptance
within the scientific community.
The system consists
of
a number
of pc boards,
each with
a
specific function,
that plug into
a standard backplane.
Table
is
a
list,
by manufacturer,
of
the various subcomponents
of
the system.
Figure
5 shows
the position of
the boards in the backplane,
and Table
gives
the assigned address
for each
I/O device.
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by rechargable Ni-Cd batteries
to eliminate transients
in ac supply,
ground
loops, and/or
60-Hz noise.
The usable system time on
a single
charge
is
shown in Figure
11.
After approximately 5.5 hours
the voltage
drops,
causing
the
current
to rise because
the supply regulator
is
set
for
constant power output.
At
an
11-volt
threshold the system shuts
off
automatically,
requiring
a manual reset
before operation can continue.
Additional electrical isolation
is provided by the use
of optical
links
from
the RNG to the
computer.
All
data are transmitted in this
manner.
Magnetic and mechanical
isolation comes from the 0.125-inch soft-
iron housing
in which all
the RNG components
are
contained.
It
is well
known that
such material provides
for attenuation of external magnetic
fields
.
An RF shield
has
also been installed to reduce
the
likelihood
of penetration by high-frequency electromagnetic disturbances.
Temperature monitoring
of
the device
is
carried out
via
a
platinum resistor.
These devices
are
considered to
be secondary tempera-
ture standards
and therefore traceable
to N.B.S.
standards.
The tempera-
ture
coefficient
of very pure platinum wire
is extremely well known and
the manufacturers
specifications
show very good stability.
The resistance
of the device
is
converted
into
a frequency that
is
in turn transmitted
to the
computer for decoding via
a third optical
link.
Figure
12 shows
that
the RNG enclosure has
sufficient
self-heating to raise the diode
to approxi-
mately 28
C over
a period of
6 hours.
The gaussian/white noise
character
of the diode output
is unaffected
by such
a
small
change.
Since we have determined that optimum diode performance occurs
at
100 pA leakage
current,
a failsafe
circuit has
been installed that
shuts
off
the RNG
if
the
device current
falls
below 95 pA or rises above
105
pA.
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PULSE
RATE
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FIGURE
LOG OF THE PULSE RATE AS MEASURED BY FREQUENCY
ANALYSIS PLOTTED AS A FUNCTION OF BAND GAP
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probability for the
carrier that
initiates
a noise pulse.
In turn,
the
temperature dependence
of the pulse rate
is dominated by the variation
of
the energy bandgap E
with temperature:
g
NoT
where
4 /2m*
E"
T = exp
-
m*
= Effective mass
of tunneling carrier
q
= Electronic charge
ft
= Planck's
constant
£ = Electric field
E
= Bandgap of silicon,
g
-1
While
this theory was
never extended to high-rate
(10
s
)
diodes such
as
the MD-20,
Figure
10 demonstrates
that agreement
of
the principal
features
of
the theory with experiment
is
still good.
The pulse rate
N
(obtained from frequency spectra) was plotted semilogrithmically as
a
3/2
3/2
function of
E
,
From the
linear dependence
of
log N on E
it
is
g
g
clear that
the variation in pulse rate with temperature
is
due to the
change
in E
with temperature,
g
From the published data on the MD-20,
the electric field
in the
breakdown volume
can be
calculated.
Assuming
all other parameters to be
well known,
the best
fit
to the
data in Figure
10 is obtained with an
effective mass
= 0,38
±
0.01 m
, which
is
of
the same order as the
o
accepted values
for effective masses
in silicon.
,
System Isolation and Monitoring
To provide
for system integrity under various environmental
conditions,
a number
of isolation steps have been taken.
Power
is supplied
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FIGURE
REPRESENTATION OF SINGLE-SWEEP PULSE TRAIN FROM NOISE
DIODE AT A CURRENT OF
150 pA
We have experimentally measured the response of unfiltered noise
pulses to variations
in reverse
current,
and have determined the mean
"dead time"
occurring randomly between sawtooth pulses
to
be:
t
= 0.01 X 10
exp[l/1.567 X 10
]
where
I
is
in amperes, and
t
is
in seconds.
The device response to other
environmental conditions was measured
by:
(1)
Subjecting the diode to an
applied magnetic field
of 6000 gauss with the
junction both perpendicular
and parallel to the
field;
and
(2)
irradiating the device with low-intensity
241
]_47
Co Y-rays
(1.33 MeV),
Am a-particles
(5.49 MeV),
and
Pm 0-radiation
(255 keV max).
As expected from the small junction area and device packag-
ing,
none
of these variations
in physical environment
caused any change in
either the pulse height
or frequency spectrum.
In the theoretical model for the behavior of
low-pulse-rate
devices,
the pulse rate
N depends
most strongly on T,
the tunneling
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PULSE
AMPLITUDE
(arbitrary
units)
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FIGURE
PULSE HEIGHT ANALYSIS OF DIODE
D-1 AT TWO LEAKAGE CURRENT VALUES
production processes
*
The narrow peak on the
left results from continuous
carrier avalanche while
the broad feature
is produced by the
large sawtooth
noise pulses.
This picture
is
supported by analysis
of
the sawtooth pulses.
At
a reverse
leakage
of
100 pA,
continuous pulse
trains are observed with
a
fast
oscilloscope.
As shown in Figure
9 considerable "dead time" in the
pulse train appears
at
a
current
of
150 pA.
As
the
current increases,
there
is
a greatly enhanced probability that
the device will stay
"on"
in the avalanche mode after
a pulse.
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FIGURE
PULSE HEIGHT DISTRIBUTION OF FILTERED DIODE NOISE FROM
-40
TO +40° C
Over
the
same temperature range,
the spectral
density
of
filtered output
from the
diode was
found to
be white noise within
±1
dB.
The unfiltered
frequency spectrum
(to
2 MHz)
showed
a shift
in the fundamental pulse rate
peak as
a function of temperature.
This
is
to be expected from the
de-
pendence
of pulse rate on bandgap energy.
Published performance
data on the MD-20 shows only that
the
spectral noise
density
is relatively constant from a reverse
current
of
60 to
120 pA and drops
off
at higher
currents.
Figure
8 shows
two pulse
height
spectra,
at
100 pA and 200 pA.
While the spectrum taken
at
100 pA
can
be easily fit with
a gaussian distribution,
it
is
clear that
at 200 pA
the spectrum is extremely asymmetric.
Since the behavior
of
this
device
at very high
currents
(~1
mA)
is
like
that
of
a purely avalanche diode— i,e.
small
noise voltage and wideband white noise—we believe the asymmetric
response shown at
a reverse current
of 200 pA
is
due to competing noise
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II
SUBSYSTEM TESTING
A.
Noise Diode RNG
.
Device
Tests
Six Texas
Instruments planar silicon noise diodes
(type MD-20)
were tested.
Data
collection was accomplished through pulse height
analysis,
frequency spectrum analysis
and
fast
signal analysis
of indi-
vidual pulse trains.
By varying such factors
as
temperature,
reverse
current,
magnetic field,
etc.,
the dependence of diode output
on each
variable was established.
The unfiltered diode output was found
to
be
random amplitude sawtooth pulses with
a rising slope of 3,1 X 10^ V/s
for
a
load capacitance
of ^10 pF and reverse
current
of
100 pA.
The
mean rate
of these pulses was
determined to be
a function of temperature
consistent with the
change
of bandgap energy.
In addition,
’’dead spaces”
were found to appear randomly between pulses.
The duration of these dead
times
depends strongly on the reverse
current.
While maintaining
a
constant reverse
current
of
100 pA,
as
measured by an electrometer,
the temperature of the diode was varied from
-40 to +40° C.
Over
this temperature range,
the pulse height
and frequency
spectra
of
several diodes were measured.
The variation
in spectral noise
density measured by plotting
the
integral of each pulse height spectrum as
a function of temperature was within a few percent
of
the specified value
of -3.2 pV//Hz
0 C
.
The
integral of each peak was also compared with the
area obtained by integrating
a gaussian curve that was
fitted to each
set
of data points.
These areas were found to agree to within
1%;
indicating
that
from -40 to +40°C,
the noise pulses maintain
a gaussian amplitude
distribution.
Figure
7 displays
the pulse height data and computer
fit,
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resolution in both the
x and
y direction
is
200 points/inch over
a 20-by-
20-inch surface.
Points
may
be entered in a stream of up to 200 points
per second or may
be entered individually.
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4)
D
Test
The D
test
consists
of
comparing the theoretical
distribution of
a random line
in 2-space with an empirical distribution
obtained from the random number generator being tested*
This
test
seeks
to determine
if there
is
a
clustering
of
numbers
in 2-space,
5)
Conditional
Bit
Test
The conditional
bit
test
is
performed to
check inde-
pendence
of bits within each number produced by
a random number generator.
Given
a
1 in
a single
bit position,
what
is
the probability that any other
bit
is also a
1.
For true
bit-by-bit
independence this
should be one-half.
C
.
Display
.
Grinnell Graphics Display System
The Grinnell Graphics Display System consists
of 256 X 512 bytes
of read/write high-speed memory,
two
sets
of three 8-bit DACs
,
and two
8 X 24 bit
color
look-up tables.
Depending upon the configuration determined
by software commands
over
a DMA channel from the LSI-11,
a single image
with high
color resolution
(8-bits
of
color /pixel)
may
be displayed on
either
or both
of two color monitors
,
or two 4-bit
color resolution images
may be displayed independently.
Because the
color
of
a given pixel
is
determined by the values
in
a
color
look-up table,
it
is possible
to
simulate animation by software manipulating
the color
of
a
complex predrawn
image
.
.
Summagraphi cs
Tablet
Graphic information in the form of an x-y coordinate point
is
presented
as
input
to
the LSI-11 over
a 16-bit parallel interface.
The
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c.
Fixed- Length Statistical Tests
1 )
Frequency Test
The frequency test analyzes
the
number of times
a
particular digit and the number of transitions between pairs
of digits
are observed
in
a random sequence.
For
the
test
of the various
sources
considered here,
a
15-bit fraction
(0 ^ fraction <
1)
is derived from the
source in question.
That fraction
is used
to calculate an integer between
1 and
10 inclusive.
A single-row matrix
is used to accumulate the transi-
tion occurrences where matrix element
(I,
J)
represents
the number
of
times
digit
I followed digit
J,
Standard
chi-square techniques are used
to determine
if there
is
no preference either among single digits
or
among paced sequences
.
2 )
Gap Test
Given
a
digit
between
0 and
derived
as
above,
the
gap test
measures
the number
of
digits
that
occur
in the sequence between
successive appearances
of
the specified digit.
The accumulated number
of
digits
in each gap are
compared to
the expected geometric distribution
by
a
chi-square
test.
This procedure
is followed for all digits
0 through
9 in the given sequence.
3)
Yule
Test
The Yule test measures
the distribution of the sum
of
consecutive decimal
digits
for the random number generator in question
and compares
the resultant distribution with
the expected
(near normal)
one
by
a
chi-square
test.
This
test
is particularly sensitive to excessively
nonuniform distribution of the digits between
0 and
9 in the sequence.
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Utilization of the above statistical procedure permits
analysis
of the binary noise sequence
for excess
l's or
0's by the most
efficient technique
currently possible.
To show this,
consider the expected number of samples
required in
a fixed-length sequence to meet the
Type
I and Type
II error
specifications,
a and
P:
n
v/po
(i
- v
- yv
- v]
:
fix
(Pi -
po
>
where
is
the z-score,
assuming
a normal distribution,
corresponding
to the
a criteria
(for a = 0,05,
z
= 1.65),
Likewise,
for
g = 0.05,
a
Z
= -1,65.
For
p
= 0.5 and
p
= 0.6,
n
= 267.
P
o
fix
The average number
of
trials required to reach
a
p^
decision and
a
p^ decision,
respectively,
is
given by
-
I
log (rfjl
*
(1 -
8)
log
Dl
~
/
p i\
P l l0g \“/
+
(1
“
P 1
)
l0g
Using the above values
for p^,
p^
,
a, and
g
,
n^
= 132.
Thus,
sequential
analysis requires approximately one-half
the number of
samples required
by the
fixed sample situation for this
case.
The general proof
is given
in Reference
3,
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(3)
If
the sum of
f s
lies
in Region
I,
do
Step
1.
(4)
If
the
sum lies
in Region
II,
stop the
run,
concluding that the binary sequence
is derived
from the undistorted p
distribution*
o
(5)
If the sum lies
in Region
III,
stop the
run,
concluding
that
the binary sequence
is
derived
from the distorted p^ distribution.
The
equations
for
the upper and
lower limit
lines shown
in Figure
6 are,
respectively:
y
=
d
+ Sn
*1
y
= -d
+
Sn
o
o
where
-
log
in which
S
is
the slope,
n
is
the
number
of samples,
and
d
and
d
are
o
the
y-axis
intercepts.
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FIGURE 6
EXAMPLE OF SEQUENTIAL SAMPLING PLOT
following
the sampling of each
bit:
continue sampling before making
a
decision
(Region
I
in Figure
6);
label the sequence
as distorted
(Region
III)
or undistorted
(Region
II).
Thus,
sampling
steps
can be
defined for the nth sample:
(1)
Sample the binary sequence
(2)
Sum the number
of
l's to date.
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higher-level
languages
such as FORTRAN and MACRO, and provides
a framework
for an extensive real-time interactive
system.
Versions
of RT-11 monitors
have been in operation for nearly ten years,
so that
this
software system
is especially trouble-free.
b.
Sequential Analysis
The sequential analysis technique
is
an extremely
efficient technique for determining whether
the output
of
a binary random
generator contains
a distribution of
0's
and
l
r s
as expected,
or is
distorted.
The principal advantage
of
the sequential sampling technique
as
compared with other methods
is
that,
on the average,
fewer bits
per
final decision are required
(roughly 50%)
for an equivalent
degree
of
reliabi lity
.
Before we are able to detect
if
the random output
of
a
binary generator
has
been distorted, we must
a priori
define
criteria
as
to how much distortion we require,
and what
statistical risks we are
willing to accept
for making an incorrect
decision.
To meet
these
criteria,
sequential analysis requires
the specification of four parameters
to
determine from which
of
two binomial distributions under
consideration
(distorted or undistorted)
a
data sample belongs.
The
four parameters
are:
p
,
the fraction
of
l
f s expected
in an undistorted distribution
(e.g.,
50%);
p
,
the fraction of
l
T s assigned
a priori
to define
a
distorted distribution
(e.g.,
60%);
a,
the
a priori assigned acceptable
probability for
concluding that
the random source
is
perturbed
(p
dis-
tribution) when in fact
it
is
not
(Type
I error);
and
(3,
the
a priori
assigned acceptable probability for
concluding that
the random source
is
unperturbed
(p
distribution) when in fact
it
is
(Type
II error).
With
the parameters
thus specified,
the sequential
sampling procedure provides
for construction of
a decision graph as shown in Figure
6.
The decision
graph gives
a procedure for making one of three possible decisions
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Table
ADDRESS ASSIGNMENTS
Device
Address
Function
Interupt
Vector
Special interface
172420
172422
172424
Command/data
Clock input
Temperature
130
Graphics tablet
167770
167774
Command/register
Data buffer
304
DRV 11-B
(Grinnell)
172410
172412
172414
Word
count
Buffer address
Command
124
ADC
172440
Command
Data
buffer
146
Functionally the system consists
of
a
16-bit
CPU,
32K words
of random access memory,
and
one
dual floppy disk drive
capable
of
double-density recording.
Special purpose
I/O interface
cards allow for
serial communication to
a CRT terminal,
16-bit
parallel
communication to
any parallel device,
and
direct memory access
to and from an external
device.
An eight-channel fully differential analog-to-digital
converter
allows the monitoring
of external analog signals.
Finally,
the system
has
a foundation module on which
a specific interface
is
constructed for
control
of the various
random sources.
Appendix B contains
the functional
description
(at
the
component
level)
of this interface.
.
Software
a.
RT-11 V03B Monitor
The software environment
of
the LSI-11 system is under
control
of DEC'S RT-11 V03B monitor.
This monitor allows for the use
of
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Table
SUBCOMPONENT MANUFACTURERS
LSI-11
Subcomponent
Function
Manufacturer
KD11-HA
CPU board
MDB Systems
MLSI-SMU
System monitor
MDB Systems
DRV- 11C
16-bit
digital
I/O
MDB Systems
MSV-09
MOS memory,
28K
MDB Systems
DT-1761
16 channel ADC
MDB Systems
DRV- 1 1
J
4 port
serial
I/O
Digital Equipment Corporation
DRV- 1 IB
DMA interface
International Data Systems
DSD-440
Floppy disk
Data
Systems Design
TCU-50
Calendar
Digital Pathways
KD11-HA
MEMORY
DEC DRV-11J
DSD-440
DRV11-B
SPECIAL-PURPOSE INTERFACE
DT-1761
DEC DRV-11C
EXPANSION SLOT
EXPANSION SLOT
MLSI-SMU
TCU-50
FIGURE
SLOT DIAGRAM—FRONT VIEW
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Table
PSEUDORANDOM SEQUENCE FROM 8-BIT SHIFT REGISTER
Sequence
Number
J
J
J+l
J+2
J+3
J+4
J+5
J+6
J+7
J+8
J+9
236
141
165
132
251
131
243
250
197
210
177
205
180
201
183
150
127
246
160
198
248
194
253
134
145
193
162
172
208
121
125
156
226
233
137
216
142
102
218
228
170
203
123
124
169
225
126
195
200
232
188
241
144
244
108
109
114
213
173
229
159
189
168
101
190
130
112
191
100
176
111
116
167
120
122
182
121
163
153
185
234
214
242
207
222
131
212
152
184
223
141
148
175
143
151
219
209
155
117
107
249
103
161
239
106
204
129
149
220
111
171
202
157
215
199
181
146
158
136
237
104
166
174
186
181
191
252
179
119
230
192
151
201
238
140
133
206
235
227
211
196
118
211
221
221
254
217
187
128
115
224
231
!
247
139
178
231
241
245
113
138
135
154
105
251
147
171
110
H
VI
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FIGURE
DIODE
TEMPERATURE
vs
TIME
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B.
ft-Decay RNG
1 .
Device Tests
Since the
147 Pm source undergoes random P-decay independent
of
external influences
such as temperature, electric field,
magnetic field,
etc., we restricted our testing to the radiation detector.
It should be
mentioned that,
in general,
measured decay rate
is
a function of both
source half-life and source-detector geometry.
In our configuration we
have minimized the
latter effect by arranging
a nearly 2 n
(steradians)
source-detector geometry.
Commercially available silicon surface barrier detectors are
quite well
characterized for noise leakage and resolution at room tempera-
ture.
Since
leakage
current
is
a function of temperature and can contribute
noise to the overall
system,
we tested the device performance over the
temperature range
20° C to
43° C.
Figure
13 shows
this
dependence.
We have
set
a
cutoff point on dc leakage
current at
~2
pA.
For this value we
can
calculate
a noise figure given by
V
= 2.35
/
e
ie
t
4q
where
V = Noise voltage
(full-width-half-maximum,
in eV)
e
= eV/pair
(3.5 for
Si)
—6
I
= Leakage
current
(amperes)
2 X 10
e
= Base
e
-6
t
= Amplifier peaking time
(seconds)
0.25 X 10
-19
q
= 1,602 X 10
coulombs
V
(FWHM)
= 19.7 keV for 2-pA leakage,
noise
Since the
low-level discriminator is
set
at
25 keV
,
this noise contribution
would not be observed,
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FIGURE
DETECTOR LEAKAGE CURRENT AS A FUNCTION OF TEMPERATURE
It
is possible that
a detector
such as
this
can undergo breakdown
resulting
in carrier avalanche noise similar to the MD-20 diode.
Should
this
occur,
the output will
consist
of the superposition of two fundamental
random processes
(i.e.,
electronic noise and 8-decay).
#
System
Isolation and Monitoring
As with the noise diode module,
protection against magnetic and
mechanical interference has been provided by
a
0,125-inch soft iron box.
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Similarly,
there
is an KF shield,
a battery power supply,
and optical
links
for data transmission.
Fail-safe provisions on the power supply are the same
as
for the
noise
diode unit.
In addition there
is an automatic shutoff should the
device
leakage
current
rise above
pA.
Quite often breakdown phenomena
of the type mentioned earlier are accompanied by a steep rise in the
leakage current.
The fail-safe point provides protection in case
of
temperature increase
as well as device breakdown.
C.
Computer Pseudorandom Generator
Extensive work
has been done
to develop various
computer algorithms
that
meet
a variety
of statistical requirements
for pseudorandom sequences.
One
of
the
best and most
studied
is
a form of Tousworth feedback shift
register generator.
In this
type
of number generation,
various
bits from
a
shift register are
summed modulo 2 and presented to the
shift register
input.
The
length of
the
shift
register and those bits used
for feedback
determine
the properties
of
the resulting sequence.
Using an algorithm by Kendall
we have adopted
a 31
bit register
with bits
31 and
13 used
as
feedback.
Reference
6 compares
a number
of
different
generators using an extensive series
of statistical
tests.
The particular shift register generator we have selected
is
considered
to
be among
the best
.
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V
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III
SYSTEM ANALYSIS
Using the random source hardware developed for this project and the
standard fixed
length statistical tests briefly described in Section
I-B,
a number
of control runs were collected and analyzed.
Five hundred
thousand
samples
in two groups
of 250,000 each were taken from each
random source and then tested with four separate procedures.
The results
are
listed in Table
4.
In addition,
the entire system was
tested for
1,000 runs
for each
source
in the
final
configuration, which utilizes
sequential analysis
in
conjunction with
the random sources.
The results are tabulated in groups
of
100 in Table
5.
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Table
RESULTS OF STATISTICAL TESTS
OF THE RANDOM NUMBER GENERATORS
Generator
Test
Digit
Bit
Chi-Square
Pseudorandom
Digit
frequency
(0-9)
9.7
(9)
Transition frequency
(all possible pairs)
80.5
(99)
Gap
12.2
(21)
18.0
17.5
24.6
17.5
17.7
24.1
23.1
23.3
22.7
Yule
100.4
(45)*
Distance
22.6
(19)
Conditional
bit
110.0
(127)
108.0
108.9
109.8
95.7
115.6
98.2
102.0
Digit
frequency
(0-9)
6.8
(9)
Transition frequency
(all possible pairs)
85.9
(99)
Gap
16.2
(21)
14.6
10.2
*
Significant results
for
p £ 0.05.
NOTE:
Numbers in parentheses
are degrees
of freedom.
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Table
(continued)
Generator
Pseudorandom
Gap
(cont’d)
(cont
f d)
Digit
Yule
Distance
Conditional
bit
Beta Decay
Digit frequency
(0-9)
Transition frequency
(all
possible
pairs)
Yule
Distance
j
Chi-Square
29.5
19.1
23.8
20.5
17.4
13.0
17.5
30.4
(45)
29.8
(19)
126.7
(172)
142.6
138.4
148.3
125.9
110.9
129.7
140.0
3.3
(9)
85.7
(99)
13.8
(21)
17.6
18.4
19.1
23.4
37.7*
21.1
13.9
21.6
18.2
33.8
(45)
17.8
(19)
Significant
results
for
p £
0,05,
NOTE:
Numbers
in parentheses
are degrees
of freedom.
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Table
(continued)
Generator
Test
Digit
Bit
Chi-Square
Beta Decay
Conditional
bit
116.6
(127)
(cont
' d)
114.6
105.4
110.9
127.6
94.2
106.5
122.3
Digit
frequency
(0-9)
7.0
(9)
Transition frequency
109.2
(99)
(all possible pairs)
Gap
18.8
(21)
31.5
15.5
22.3
19.5
22.1
15.4
20.8
31.3
20.4
Yule
41.5
(45)
Distance
18.3
(19)
Conditional
bit
124.4
(127)
118.1
116.7
126.1
110.5
123.2
105.3
133.4
*
Significant results
for
p £
0,05.
NOTE:
Numbers
in parentheses
are degrees
of
freedom.
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Table
(continued)
Generator
Test
Digit
Bit
Chi-Square
Noise Diode
Digit
frequency
(0-9)
8.1
(9)
Transition frequency
(all possible pairs)
100.6
(99)
Gap
21.0
(21)
19.7
25.3
21.2
27.5
10.2
24.2
9.5
31.4
12.8
Yule
32.8
(45)
Distance
24.0
(19)
Conditional
bit
132.2
(127)
153.6*
135.0
121.3
135.5
132.1
143.9
136 .
Digit frequency
(0-9)
10.0
(9)
Transition frequency
(all possible pairs)
85.3
(99)
Gap
23.9
(21)
18.0
18.6
17.7
20.9
12.5
*
Significant results
for p £
0.05.
NOTE:
Numbers
in parentheses
are
degrees
of freedom.
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Table
(concluded)
Generator
Test
Digit
Bit
Chi-Square
Noide Diode
Gap
(cont’d)
10.3
(cont
T d)
16.4
23.2
25.0
Yule
42.7
(45)
Distance
16.9
(19)
Conditional
bit
142.7
(127)
128.3
109.6
121.4
125.8
155.6*
127.2
127.8
*
Significant
results
for
p ^ 0.05,
NOTE:
Numbers
in parentheses
are
degrees
of
freedom.
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Table
SEQUENTIAL ANALYSIS OF RANDOM SOURCES
IN SETS OF
100 TRIALS
Set
Number
P-Decay
Diode
Pseudorandom
2*
Significant with
p <
.001.
NOTE:
Table entries
are
the number of trials
out
of
100 that
met
the sequential
analytic decision criterias.
The
expected value
is
10.
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IV
CONCLUSIONS
We have shown by extensive environmental
testing and Monte Carlo
techniques
that
the LSI-11
based random number generator and analyzer
meets
all appropriate
criteria
for accepted definitions
of randomness.
Specifically,
approximately
1C)
6 bits were analyzed in exactly the
same
conditions
that will
be utilized during
the experimental phase
of
this
program
.
These
tests
showed no unexpected deviations
from the appropriate
chance
distributions.
Environmental and electrical
isolations
of
the sources
and manual
reset
fail-safe
circuitry assures
reliable operation.
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Appendix A
TECHNICAL CONSTRUCTION DETAILS OF RANDOM SOURCES
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Appendix A
TECHNICAL CONSTRUCTION DETAILS OF RANDOM SOURCES
.
Introduction
The
need for,
and the requirements
to be met
by,
a
noise generator
system have been established and described elsewhere.
In quick review,
the performance specifications and goals
call
for
a stable generation
source
of white noise
that
is protected from external physical influence
or
its
susceptibility to those effects
is known in advance.
That noise
then needs
to be
formatted in
a
specific way and transmitted
to a remote
processing unit,
again with minimal disturbance by the external environ-
ment,
The ultimate goal
is
a statistically stable random signal that
can
be measured over
long
periods
of
time with no variation with environmental
changes
.
Two sources
for the random signal were
chosen.
First,
a special
avalanche breakdown mode
(zener)
diode was
selected that produces
a very
predictable
series
of quasi-impulses essentially random over
a reasonable
time range and amplitude range.
Although external circuit
component
values
affect
the spectral
density function,
and
the applied reverse bias voltage
affects
the avalanche breakdown
levels,
those
conditions
can be controlled
closely and hence their effects
reduced to an acceptable
level.
Temperature
also affects noise statistics
but again in
a very predictable and stable
fashion,
so
a measure
of
diode temperature
is necessary during the operational
usage.
The second source
of random noise
is obtained from a radioactive
material
of known radiation intensity and statistics.
For all practical
purposes
it emits
only
low energy electrons
(225 keV max),
so
a silicon
surface barrier detector can be used to generate an electrical signal
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proportional
to the energy of the incoming radiation.
Except for the
detector,
the phenomenon is nearly independent
of
the
local environment.
Packaging of
the
sources and sensors
along with the associated
electronic circuitry provides
a
first
level
of protection from external
influences
such
as
heat,
light, vibration,
and electromagnetic radiation.
To minimize
the possibility of external electrical influence, the source
of power
for
the units
is provided by internal
chemical batteries.
To
avoid
coupling
of extraneous electrical signals
from either the remote
processing unit
or nearby equipment,
the
data
link from the noise source
unit
is provided entirely by use
of fiber optic
cables.
The unit
case
is
fabricated
from mild steel
that
provides
shielding from both magnetic
and electric fields.
.
Noise Diode
Source
The
heart
of
this unit
is
the avalanche zener diode.
It produces
a
noise-like voltage when reverse biased with
a
constant
current
dc source
of about
100 pA.
The average zener breakdown occurs
at
about
12 V and
the noise voltage
spikes
ramp continue
for
a
few volts about
that
level.
a
.
Bias
Circuit
For simplicity and practical realization,
the diode bias
current
is provided by
a
67.5-Vdc battery and
a
series
dropping resistor to a
shunt 30-V zener diode for voltage regulation,
and then on through another
resistor
of
the appropriate value
to the
noise diode.
A large bypass
capacitor
is used across
the regulator zener diode
to keep incidental
noise from reaching the
noise diode.
Since
this
is really not
a
constant
current
source,
the diode
current,
and hence
its operating condition,
can
change.
It
has been
found that
the noise statistic varies significantly
if
the
bias
current
changes more than about plus
or minus
5%,
and so
it
was necessary to include
a
capability to monitor
that
bias
current.
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b
.
Bias
Current Monitor
An operational amplifier was employed
as
a trans conductance
amplifier
to
convert
the bias
current to
a voltage measure and to keep
the diode
at
a virtual
ground.
The feedback resistor was
chosen to
be
k0, which then provides
a
-5 Vdc output.
An inverting amplifier with
unit
gain generates
a +5 Vdc output
that
is used for comparison against
a
limit window
of
4.75 to
5.25
Vdc.
If
the bias
current varies
more than
±5% of
nominal
,
then the comparator test
circuit
output
can be used
for
warning or
control.
c
.
Noise
Preamplifier
The
output
of
the
noise
diode
is
a voltage ranging from about
V upward
in sawtooth fashion to
14 to
V and with an extremely fast
fall
time back to
V.
The
diode
chip capacitance and the
stray capaci-
tance
of
the wire
leads establish
the minimum, which
is about
or
pF
.
The
charging resistance
is essentially the bias
supply resistor value
of
180 k^ and hence
it
is necessary to use an amplifier with
a very
low
input
capacitance and
a quite high input resistance
if
the noise voltage
waveform,
and hence
its
spectral response,
is
not
to be influenced greatly.
This
is achieved
by use
of
a National LM310N unity gain buffer amplifier
which has
a specified input
capacitance
of
only
1.5 pF,
an input resistance
of
1010 ohms,
and
a slew-rate
of
at
least
30 V/ps
, which maintains
the
desired system bandwidth,
d.
Noise Bandpass Filter
Although
the noise diode power frequency spectrum continues
to
at
least
5 MHz,
only that portion from
1 kHz
to 200 kHz
is
desired for
use
by the processing unit.
Thus,
a
cascade
of
a high-pass
and
a low-pass
active filter
is used.
Both filters
contain three poles
so
as
to provide
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dB per octave rolloff,
and the design is
a Butterworth for smooth
amplitude and phase response.
Again,
the National
1AI308A voltage follower
is employed
for the active filter elements.
e .
Noise Output
Two outputs
to the processor unit are provided.
The
first
is
the bandpass
filter output time waveform in analog form,
and the second
is
a zero-crossing digital pulse train.
1 )
Analog Output
The analog output
is provided by use
of
a video bandwidth
fiber-optic data
link.
Because the peak voltage from the bandpass filter
is
less than
100 mV,
a post-amplifier
is necessary as well
as
a current
booster to drive the
light emitting diode
circuitry.
A National LF357N
FET operational amplifier and
a LH0002CH current booster amplifier are
configured as
an X10 gain noninverting signal amplifier with adequate
bandwidth and linearity to faithfully reproduce the noise signal.
A
Meret
,
Inc.,
analog optic transmitter composed
of
a current
linearizer
and the LED then converts
the analog noise signal
to an optical replica
for transmission over
a fiber optic
cable.
2)
Digital Output
The zero-crossing digital representation of
the noise
is
provided by first
a standard analog voltage comparator
(a National LM2903N)
followed by a CMOS one-shot multivibrator
chip
(Motorola MC14528BP)
, which
establishes
the minimum pulsewidth at
5 ps
,
and then a
single transistor
(2N2222 )
to drive the Meret,
Inc.
digital optic transmitter
LED.
Thus,
each time the noise voltage from the fiber passes through zero in the
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positive-going direction,
a
ps optical
pulse
is applied to the digital
fiber optic cable.
f
•
Test and Shutdown
Sxnce the noise unit
is powered by internal batteries
of limited
power capability,
it
is
necessary to monitor the state of the battery and
shut
down the optic
links
if
that voltage drops
below a
level that
is
known to cause erroneous
operation.
A simple analog voltage comparator
(LM2903N)
is used
to sense when
the battery voltage
drops
below that
level,
and
the output
of
the comparator
is used to inhibit
the one-shot multi-
vibrator
in the digital optic data
link.
Thus,
the operator at
the data
processing
center
can detect
a
low-battery condition if the digital
link
ceases
to
transmit
but
the analog continues
to function.
Similarily,
if
the noise diode bias
current
varies outside the ±5% of nominal window,
the
comparator output
is used to shut down the digital optic data
link.
of these
fail
signals will toggle
a simple
latch circuit
(CA4011B),
which then shuts
off
the digital optic data
link.
The
latch
is reset each
time
the unit
is
powered down and then turned on again.
There
is
about
a
10-s
time delay to allow for the unit
to reach normal operation before
the
latch
is
released.
£
•
Temperature Monitor
Since
the noise
diode output
is temperature-sensitive,
a measure
of the diode holder temperature
is made and passed out to the data processor
via
a
low-speed fiber optic data
link.
A platinum resistance thermometer
element
is
imbedded
in the
diode holder, and
a precision resistance bridge
circuit
feeds
the temperature difference voltage to a precision instru-
mentation amplifier
(BB3630AM), where it
is
scaled to 100 mV per degree
Celsius.
A voltage-to-frequency- converter
(BB-VCC42 ) then converts
the
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signal
to
100 Hz per degree Celcius.
This
digital frequency pulse stream
is then applied to
a digital optic transmitter
(BB3713T)
that converts
the
pulse train to an optic replica for transmission over
a fiber optic cable.
The accuracy of the system
is
0.5 C from 10 to
C.
h
.
Power Distribution
Several different voltages are needed for the
circuit elements.
First,
the noise diode
is operated from +30 Vdc, which
is
not shared by
any other circuitry to avoid interaction effects.
The noise amplifiers
and active filters
are supplied with
a +10 Vdc while
the peripheral
amplifiers
are provided with ±12 Vdc.
The digital optic transmitter LED
uses
+5 Vdc, while
the analog optic transmitter LED and current
linearizer
use +12 Vdc.
A special +10 Vdc source
is used by the temperature-sensor
bridge
circuit.
Finally,
the direct battery voltage
at
a nominal
14 Vdc
is used for the
low battery test
circuit.
All
of these voltage
lines
are
adequately bypassed using
large value tantalum capacitors with decoupling
ceramic capacitors
located at
each integrated circuit.
i
.
Circuit Layout
All
of
the
circuitry is
located on
a single printed circuit
board that was especially designed for analog hand-wire layout and assembly.
Large-area ground-plane
is provided so as
to minimize
coupling effects
from either adjacent
or remote circuitry.
Componeijt
lead-length
is kept
to
a minimum and inter connective wires are run point-to-point along the
ground plane.
Parallel bundling
of wires
is avoided.
Power supply
voltages are attached
at one end of the
PCB with direct bypassing.
Signal
flow layout
is approximately linear down the board to the opposite end
where the optic transmitters
(high current
loads) are located.
All discrete
components and integrated circuits are located on one side
of
the
PCB, while
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the majority of the interconnective wiring
is
on the opposite side*
PCB
mechanical support
is provided by four standoff posts,
one
at
each corner
electrically tied
to the PCB ground plane*
j
•
Noise Diode Mechanical Mount
The
noise
diode and the thermometer are thermally coupled with
a
low temperature
solder inside
a small
cylinder of
stainless
steel.
The
electrical
leads were
first painted with an insulating varnish and then
held
in place with epoxy resin.
This
type
of mount provides
a firm
mechanical support
and
a
good thermal
link between the diode
case and the
thermometer element.
The holder
is
then attached
to the PCB using
a metal
clamp and screw, with the whole structure acting
as
a grounded electric
shield.
The
diode and thermometer electrical
leads
exit from the ends
of
the holder
cylinder.
.
Electron Noise Source
This
unit
derives
its random noise from
a radioactive material
that
emits electrons
of
a random energy below 225 keV at random time intervals.
These electrons
are detected by use
of
a
large-area silicon diode
that
is
reverse-biased
at
100 Vdc to establish the
proper
charge collection volume,
a
.
Bias
Circuitry
The radiation detector diode operates
best
for this application
when reverse biased at about
100 Vdc from a voltage source impedance
as
high as
practical.
At that
bias voltage,
the detector has
a small
but
significant
leakage
current
of about 300 nA, which therefore requires
that
the source resistance
be
less than about
10 M0 so as
not
to cause
a
significant
bias
voltage drop.
At 300 nA and
10 MO,
the drop
is only
V,
which
is acceptable.
Thus,
the
bias
circuit
is
a +100 Vdc supply,
a
1 MO
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series resistor,
and a shunt
0.1 pF capacitor for filtering purposes
followed by the
llQ detector
load resistor and then the detector itself.
b #
Leakage Current Monitor
Since
the detector
leakage
current is
a sensitive measure
of
device integrity and
can contribute noise to the electron signal,
this
current
is measured and used to shut down the system if
leakage exceeds
a
predetermined
limit.
As was
done in the noise diode source,
a transcon-
ductance amplifier
(LF 355 N)
is used in the
common or ground
lead of the
detector.
It maintains
the anode at
a virtual ground along with large
capacitance shunting as well
as providing an output voltage proportional
to leakage current.
The
feedback resistor value was
selected
as
Mfi,
which scales
the
current to -3 V per
pA.
A unity-gain inverting amplifier
(LM 741
V) then applies the voltage to
a comparator
(LM 2903
N)
that
has
a threshold of +6 Vdc.
Thus,
a shutdown command
is generated if the
leakage current exceeds
2 pA--a level where the
noise contribution to the
electron signal
is
still below the discriminator threshold.
c.
Noise Preamplifier
This
type
of noise
is
best processed by use of a charge-sensitive
amplifier followed by a double differentiation to extract only the very
fast
diode discharges when an electron passes through
it and to ignore the
slow
"tail" recharge.
This
is performed by use
of
a commercial hybrid
unit
(AMPTEK A-203).
At
the output
of this device there exists
a series
of random height
"impulses" that occur at random times,
each impulse
corresponding to a single electron that passed through the diode.
d.
Post Amplifiers
The output
of the
charge-sensitive amplifier
is
quite small
in
amplitude, and
so
it
is
necessary to amplify
it further before it
can be
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used.
This amplification
is provided by a
chain of three
IC amplifiers.
First,
a single
low-level,
wi de-bandwidth operational amplifier
(NE531V)
increases
the signal
by 10 from about
10 mV up to about
100 mV.
This
is
followed by a two-state high-level, wi de-bandwidth amplifier composed of
an op amp
(LF357N)
and
a
current amplifier
(LM0002CH) which provides another
times-10 gam.
At
that
point,
the noise pulses are about
1 V in amplitude
and
can be processed
for transmission.
e
•
Data Transmission
Two forms
of the noise signal are generated and transmitted to
the remote processor unit.
The first
is an analog replica of the noise
while the second
is
a digital pulse replica from a threshold
level
discriminator
,
1 )
Analog Output
The
entire analog voltage waveform from the post amplifier
is used without bandlimiting to drive the MERET analog optic transmitter
LED via
the
current
linearizer.
Since the optic
link has
a bandwidth of
at
least
5 MHz, essentially all
of
the analog noise signal spectrum is
transmitted without
significant degradation.
2)
Digital Output
The same analog signal
is then applied
to
a threshold-level
discriminator
(AMPTEK A-206)
for
conversion to pulse form.
The amplitude
threshold is
set
for
a
lower-level pulse energy output of 25 keV,
and the
resulting signal
is
then applied
to the digital optic transmitter LED
through
a standard
IC TTL Gate
(SN74LS00).
The discriminator also includes
a minimum-width one-shot multivibrator that
sets
the digital pulsewidth
to
ps
.
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f
.
Test
and Shutdown
Three operating parameters are monitored and,
if out
of
limit,
are used to shut
down the digital optic data
link.
The
first
is
the main
system dc power
in
the form of the battery voltage.
A minimum voltage
of
12 Vdc
is
the threshold for
the nominal
14-V battery voltage and
corresponds
to the condition where about 95% of
the available energy
is
consumed.
When the
12-V threshold
is
reached,
a
comparator
(LM2903N)
senses
it and
shuts
down the
optic
data
link via
the TTL AND Gate.
The
second parameter
is
the power supply voltage used for the
bias supply for
the
electi’on detector
diode.
When that voltage falls
to +110 Vdc from
its
initial value
of +135 Vdc,
another comparator senses
that
condition
and shuts
off the
data
link.
Finally, when the electron detector diode
leakage
current
exceeds
pA,
a third comparator
senses that
condition
and also
shuts
down the
data
link.
Thus,
any normal
failure condition
will
be sensed and used to
shut
down the
digital
data
link
as
a warning
to
the
processor.
Any one
of the three failure signals will
toggle
a
simple
latch
circuit
(CA4011B), which
in turn shuts
off
the digital optic
data
link.
The latch
is
reset whenever the unit
turned off and then turned
on again.
There
is
about
a
10-s
time delay to allow for the
unit
to reach
normal operation before
the
latch
is
released.
g
.
Power Distribution
The various power supply voltages required by the
circuitry are
brought
into the
circuit
board, bypassed using
large-value tantalum
capacitors,
and then distributed
to
the
circuit points.
Additional
capacitive bypassing and decoupling are provided at each critical circuit
point
and integrated circuit.
Sensitive portions
of the circuit
are
supplied with isolated power voltages
such as
the detector
diode
, and the
signal amplifier
chain.
The high-power amplifiers and the optic transmitter
LEDs
are powered separately to avoid feedback to
the
low-level sections.
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h.
Circuit Layout
Layout
of the electron noise source PCB
is essentially the
same
as
that used
for the noise diode source
(see Section 2-h).
•
Electron Source and Detector Mechanical Mount
The radiation source
is
a
commercial unit
in the form of
a thin
metal
disk about
1 inch in diameter and
0.1 inch thick.
The electron
detector diode
is also
a commercial package
in the form of
a
1-inch-
diameter
cylinder about
0.7 inch
long with
a
BNC type of
connector on
the end opposite the diode sensitive surface.
These two units
are held
together with the radiation source next
to the diode
surface by use
of
a specially made plastic holder.
The holder
is then mounted to the
circuit
PCB such
that
a standard BNC barrel
connector and
a bulkhead connector tie
the detector to the
PCB both electrically and mechanically.
The radiation
source
can be removed easily through
a
top cover that
is
held on by
a
metal
spring
clip.
When the units
are
in place,
they form an electrically
conductive shell
that also provides
closure to visible
light.
This
is
necessary because
the radiation detector
is
quite sensitive
to photons
as well
as electrons.
4 .
Power System
Since both the noise diode and the electron noise sources require
essentially the
same dc power supply voltages,
a common power source
and regulator circuit
were developed.
The major difference
is
in the
need
of
a higher voltage for the electron detector than that needed for
the
noise diode.
Power consumption is higher for the noise diode source
since
it includes
the temperature monitor and data transmitter and
a few
more
ICs
that are
not needed for the electron noise source.
The main
power source
is
a
12-Vdc NICAD rechargeable battery, while the bias supply
is
a
set
of carbon-zinc dry-cell
batteries.
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a .
Batteries
Most
of the power
is provided by two
6—V NICAD battery packs
series— connected to provide
12 Vdc.
The desired operating period per
day was two 3-hour runs and the total
current
drain on the battery is
210 mA for the noise diode and about
160 mA for
the electron noise source.
This requires
that the battery have
at
least
a
1.3 ampere-hour capacity
when discharged at 210 mA for
6 hours.
The batteries
chosen
(Eveready N91)
have
a
1.5 ampere-hour
capacity and hence can
just
meet
the requirements
if they are fully charged.
Charging
of this
type of battery
is achieved
after
14 hours
at
150 mA and hence can be ready each day after an overnight
charging.
The batteries
are sealed units
and
so present
no hazard to
either equipment
or personnel.
The bias voltage
for each
noise
source
is provided by independent
carbon-zinc dry
cells.
Since the current
drain is
quite small,
these
types
of batteries provide hundreds
of hours
of operation before
cell
voltage drops
below the useful
level.
The noise
diode requires
30 Vdc
at
100 pA, which
is provided by
a single 45-V battery
(Eveready #415).
The
45-V battery voltage
is
dropped and regulated to 30 Vdc by use
of
a
series
resistor and
a shunt zener diode followed by
a L-C low—pass
filter.
The electron detector diode requires
100 Vdc at about
pA, which is
provided by two 67.5-V batteries
connected in series for a total
of
135 Vdc.
This
is
dropped and regulated to +100 V using a series resistor and
a
shunt zener
diode also followed by
a L-C low-pass
filter.
b
.
DC Voltage Regulators
Several dc voltages
are needed by the electronic circuitry,
integrated circuits,
and optic transmitters.
Much
of
the
current
is
required
at
+12 Vdc,
and
so the
14-V battery voltage
is reduced to +12 V
by use
of
a special series-pass
regulator
circuit.
This circuit has
the
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distinct
characteristic
of regulating
to within
0.3 V of the battery
voltage,
a feature
not normally available in
IC regulators.
Thus,
the
battery voltage
can fall
to
12,3 V before any of the
circuit power voltages
fall
below
12.0
V.
Two +10 Vdc supplies are needed,
one by the low-level sensitive
signal amplifiers,
and one by the temperature sensor bridge
circuit.
These are provided by use
of standard
IC regulators
(LM340LAH-10)
.
A
single +5
V power supply
is required by both TTL logic
ICs and the digital
optic transmitter LEDs.
This
is
provided
by use of another
IC regulator
(LM340LAH-5 )
.
The
negative
12 V needed by various amplifiers
is generated
in a
small hybrid switching
dc-dc converter
(W5R-12).
All
of
these power supply regulators are followed by L-C low-
pass
filters using either powdered iron core
or ferrite pot
cores for
high inductance
at
the static dc current.
Large-value tantalum capacitors
provide the
low shunt
impedance
to ground.
Smaller RF types
of feed-
through capacitors
are employed
to help reduce any RF pickup that
may
exist
in the power
system.
c
.
Interconnection
A double-pole,
double-throw switch either connects
the main
battery and the
dry cells to the circuitry or disconnects them and
connects
the NICAD battery to a battery charger
jack.
A 1/4-A fuse
protects
the battery against
inadvertent
internal short
circuit.
5 .
Packagi ng
Both the noise diode and the electron noise sources
are mounted in
enclosures
of identical dimensions.
The noise diode unit
has an extra
aperture for the temperature optic transmitter cable
connection.
The
internal
layout
of each
is
the same except for the minor positioning of
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v
the
carbon-zinc dry
cell batteries,
one for the noise diode unit and two
for the electron noise unit.
The enclosure
is
fabricated from mild steel
so
as
to provide both
electric and magnetic shielding for the contained circuitry.
All
seams
are welded and the
cover has
an overlapping flange.
Six
screws
attach
the
cover and
a special
RF gasket
seals
the enclosure along
the mating
surfaces.
Internally,
there
is
a
steel partition separating the power
supply compartment
from the electronics
compartment
so
as
to minimize
any interaction between the two.
The power supply wires
pass
through
the partition via
RF type feedthrough capacitors.
The batteries are held
in place by conformal foam rubber and stryo-
foam plastic.
The power supply regulator
circuitry
is mounted on
a
Vector composition board, which
is
then attached to the partition using
stand-off
posts
and machine
screws.
The on-off power switch,
the fuse
holder,
and the battery charger receptacle are mounted on the
side wall
of
the enclosure.
The
noise
source
circuit
board
is mounted
in the other part
of
the
enclosure
on stand-off
posts
and machine
screws
.
The
three optic trans-
mitters are attached to the circuit
PCB and
the alignment
is
such that
the fiber-optic cable connectors protrude through openings
in the end
wall
of
the enclosure.
Although this configuration does
not provide
the
best
of RF shielding
,
it
was
chosen as
a reasonable alternative
to
ease
the problem of electrical
connection.
A set
of
short,
insulated
wires
connect the
PCB to the partition feedthrough capacitors
for
the
various
dc voltages.
By simply removing the
cover and the four mounting
screws,
the
circuit
PCB may be
lifted out while still connected electrically
for diagnostic testing and repair.
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j*
.
Data Receiver Unit
The analog and digital signals
that are generated in the two noise
units
are transmitted to the processor unit
by use
of
fiber-optic cables.
The noise units
contain the optic transmitters,
so
a special receiver
must
be provided to reconstitute the electrical signals.
This function
is achieved
in
a single package incorporating all
of
the matching optic
receivers,
power supplies,
and signal
line
drivers.
a
.
Analog Receivers
These units
are special linear optic receivers with wide bandwidth
(5 MHz)
and good dynamic range.
They reconstitute
the analog signal
as
transmitted from the noise units and provide
it
at
a
level
of about
1 V
peak-to-peak
.
Since
the
output
level and impedance are
not
compatible
with
the requirements
of
the processor analog-to-digi tal
converter
(ADC),
a
signal amplifier and
line
driver are provided.
This
is
a single wide-
band op-amp that provides
a
voltage gain of X5 and an output impedance
of about
100 ohms.
Thus,
the signal
is available at
a
level
of about
V peak-to-peak
at
a
low
source impedance.
Several
feet
of 50-ohm cable
can be driven without
significant bandwidth or
level
loss.
b,
Digital Receivers
These units are also special optic receivers matched to the
transmitter
characteristics
and can handle standard TTL bit streams
of
at
a pulse rate
of
several megahertz.
No special
post amplifiers are
needed except
for
a simple TTL buffer to isolate the receiver to inadvertent
external high voltages.
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c.
Auxilary Analog Comparators
Two special analog comparator units with adjustable threshold
voltages are
included.
These
can accept
input
signals
of
as much as
plus and minus
12 V peak excursions
and with bandwidths
of
several mega-
hertz,
The
comparator threshold
is adjustable
over
a
±4 V range by means
of
a
front
panel
1-turn potentiometer.
The output
of
the
comparator
triggers
a one-shot multivibrator that produces
a TTL output pulse with
a width
of about
0.1
ps
.
Hysteresis
(100 mV)
is provided
in the comparator
input
circuitry
to suppress noise-induced oscillations.
d
.
Power Supply
The optic receivers and the auxilary amplifiers,
comparators,
and one-shots require
-12 Vdc and +5 Vdc reasonably regulated and filtered.
A simple,
special-purpose triple
supply using
commercial
IC regulators
powered
from
a
115 Vac full-wave rectified transformer
is
provided.
It
is
constructed on an aluminum shelf that
also serves
as
the regulator
heat
sink.
e
.
Packaging
A standard
7 ,5-inch-high-by-19-inch-rack panel supports
the
optic receiver modules,
the peripheral
circuitry,
and the power supply
shelf.
Since the optic receiver modules have BNC output
connectors,
a
simple
clamp and connector arrangement hold them to
the
front
of
the panel
so that the fiber-optic cable
can be attached easily.
The
cables
hang
vertically down for minimum strain,
since they are
somewhat fragile.
The
final outputs
of the five optic receivers
are provided via
bulkhead BNC connectors
at the top of
the panel directly in line with
each optic module.
Thus,
connection to the processor interface
is
by
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%
REFERENCES
1.
R.
H.
Haitz,
"Controlled Noise Generation with Avalanche Diodes
I.
Low Pulse
Rate Design,"
IEEE Trans,
on Electron Devices
,
Vol
.
(April
1965).
2.
R,
H.
Haitz,
"Controlled Noise Generation with Avalanche Diodes
II.
High Pulse Rate Design,"
IEEE Trans,
on Electron Devices
,
Vol.
(March
1966).
3.
A.
Wald,
Sequential Analysis
(Dover Publications,
Inc.,
New York,
N.Y.,
1973).
4.
M.
Fisz
,
Probability Theory and Mathematical
Statistics
,
pp.
584-611
(John Wiley & Sons,
Inc,,
New York,
N.Y.
,
1973).
5.
S,
W,
Golomb,
Shift Register Sequences
(Holden-Day,
Inc.,
San
Francisco,
CA
,
1967).
6.
T.
G,
Lewis,
Distribution Sampling for Computer Simulation
(Lexington Books,
Lexington,
MA
,
1975).
7.
R.
H.
Haitz,
"Mechanisms Contributing
to the Noise Pulse Rate
of
Avalanche Diodes,"
J.
Appl
.
Phys
.
,
Vol,
36,
No.
10,
pp.
3123-3131
(1965).
_
~
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