Patent GB-947810-A (GB)
Title: High resolution scanning radar
IPC: G01S13/44
Filed: 1960-07-29
Abstract: 947,810. Aerials; pulse radar. NORTH AMERICAN AVIATION Inc. July 29, 1960 [Aug. 6, 1959], No. 26604/60. Headings H4A and H4D. A vehicle borne radar system comprises a pulsed antenna for transmitting pulses of a selected frequency and receiving reflected pulses shifted by target doppler frequency; means for generating a synthetic antenna as said system moves in a selected direction including a reference signal generator defining the axis of said synthetic antenna, means for comparing said reflected pulses with the reference frequency signal to generate a coherent signal of a frequency in accordance with the difference between said reference frequency and the doppler frequency of the reflected pulses, and means for integrating said coherent signal to produce a high resolution output signal for reflections from targets in a direction determined by said reference frequency ; means for causing said antenna to scan, means for comparing the direction of said antenna with the direction of said synthetic axis, and means responsive to said comparing means for varying said reference frequency in a sense to maintain alignment of said synthetic axis with the direction of said antenna. The object of the invention is to provide a high resolution scanning radar for use in aircraft employing a physical antenna of small dimensions. In one example described the beamwidth of the synthetic antenna is 0.2 degrees and the length of the physical antenna is 30 inches. As shown, the transmitting portion of a monopulse radar system, Fig. 4, comprises a physical antenna 30 scanned at least in azimuth by a servo 31 controlled by a programmer 32, a pick-off 33 providing a signal indicative of the instantaneous position of the antenna boresight axis. The outputs of a local oscillator 34 and I.F. oscillator 35 are combined in a single sideband modulator 36 having its output fed to a transmitter modulator 37 synchronized from the radar system trigger to produce a train of pulses of the upper sideband frequency at a pulse repetition rate determined by the trigger. The pulses from modulator 37 are fed to the antenna 30 through sum and error channel duplexers 38, 38a and a monopulse bridge 39, echoes received by the antenna being fed to the sum and error channels of the receiver via leads 40, 41. The sum signal is large for a target within the physical beam while the azimuth error or difference signal is zero for a target in the boresight axis and increases in proportion to azimuth angular distance off axis. The error signal is in phase with the sum signal for targets on one side of the axis and out of phase by 180 degrees for targets on the other side of the axis. Fig. 5 shows the receiver. Considering the sum channel, the echo received from a target is shifted with respect to the transmitted frequency by the target doppler frequency fd, this signal being fed to a sum channel mixer 50 also having an input from L.O. 34 to remove the local oscillator frequency. The signal from mixer 50 is fed through I.F. amplifier 51 to a pair of phase detectors 52, 53. A coherent I.F. reference signal for the detectors is generated in a single sideband modulator 54 by combining therein the I.F. signal from oscillator 35 with a reference frequency f<SP>1</SP>d from a variable frequency oscillator 55. Detector 53 handles the in-phase component and is directly coupled to modulator 54 whilst detector 52 which handles the quadrature component is coupled to the modulator via a 90 degrees phase shifter 56. The output of each phase detector 52, 53 comprises the coherent video such as signal 24, Fig. 7, which varies sinusoidally at the frequency fd-f<SP>1</SP>d, the video signals being fed to an integrator 57 of the magnetic disc recorder type (described in detail with reference to Figs. 6a, 6b, not shown) including a storage device 58 and a summing device 59. The integrator stores a number of successive video pulses (e.g. 25 pulses) and adds them algebraically. For a target lying in such a direction that its doppler frequency fd differs appreciably from the frequency f<SP>1</SP>d, the signal 24 alternates at the difference frequency during the integration period resulting in a zero output from the integrator. However, when the frequency difference is substantially zero the integrator produces an output which is rectified at 60 to obtain a signal proportional to target amplitude, this signal being applied to a conventional cathode-ray tube indicator 61 having a sweep circuit 85 controlled by the pick-off 33. Thus as the axis of the synthetic antenna is scanned by varying the frequency f<SP>1</SP>d only targets lying in the very narrow beamwidth of the synthetic antenna produce a signal from the integrator 57. The axis of the synthetic antenna is slaved to the boresight axis of the physical antenna 30 by utilizing the signal from the error channel duplexer 38a and causing it to vary the oscillator 55. The signal from lead 41 is fed to a mixer 70 having an input from L.O. 34, thence to I.F. amplifier 71 and to error channel quadrature and in-phase phase detectors 72, 73 which are provided with the quadrature and the inphase components of the reference frequency f<SP>1</SP>d from the phase shifter 56 and modulator 54. The outputs of the detectors 72, 73 are fed to the storage device 58 and combined in a summing network 74, the outputs of the networks 59, 74 being fed to a phase detector 75 producing a signal indicative of the magnitude and sense of the error channel signal with respect to the sum signal. The output of detector 75 is fed through a gate 76 controlled by a generator 77 synchronized from the radar system trigger, the gated error signal being fed through filter 78 and amplifier 79 as a control signal to effect variation of the frequency of oscillator 55. Thus the frequency f<SP>1</SP>d is varied to null the error signal from detector 75 and make the synthetic axis coincide with the boresight axis. Fig. 12a comprises a plot of the gain of the sum channel of the physical antenna against the angle of the target off the boresight axis (#- #A), where #A is the direction of the boresight axis and # is the target direction. Fig. 12b comprises a similar graph of the antenna gain for the error channel whilst Fig. 12c shows the gain of the synthetic antenna as a function of the angle of the target off the synthetic axis #o. The net response to a target is the product of the gains of the synthetic and physical antenna for each of the two channels and Figs. 13a-13f show the composite gains, Figs. 13a, 13c, 13e being related to the sum channel and Figs. 13b, 13d, 13f to the error channel. Figs. 13c, 13d show the gain when #A=0O, i.e. the physical and synthetic axes are aligned, Figs. 13a, 13b, when #A is smaller than #O, i.e. the physical antenna is to the left of the synthetic antenna and Figs. 13e, 13f when #A is greater than #O, i.e. the physical antenna is to the right of the synthetic antenna. To ensure that the error signal from phase detector 75 is proportional only to angular error and not to reflective strength of the target, the gain of the I.F. amplifiers 51, 71 may be controlled from the sum signal to hold the average output of the sum amplifier 51 constant over the period of a selected gating interval. Thus the output of amplifier 51 is fed to a detector 90 and thence via gate 91 to the automatic gain control amplifier 92 controlling the two amplifiers 51, 71. Gate 91 is also controlled from the gate generator 77 which produces a relatively long gate (i.e. 10 miles or more of equivalent range) suitably delayed to centre the gating interval about the range of greatest interest. On such a long range interval there will be approximately an equal preponderance of targets on both sides of the synthetic axis and thus the average error voltage will be proportional to #A-#O. The actual dimensions and characteristics of such a radar system are calculated in the Specification.