Patent GB-1431147-A (GB)
Title: Signal generator for testing vor navigation receivers
IPC: H03H17/08
Filed: 1974-05-21
Abstract: 1431147 Signal generator for testing radionavigation receivers; AFC systems BENDIX CORP 21 May 1974 [12 June 1973] 22726/74 Headings H4D and H3A A signal generator for testing a VOR receiver, of the kind which provides bearing information by comparing the directionally-variable phase of one signal with the fixed reference phase of another signal, comprises a stable oscillator operating at a frequency which is an harmonic of the frequency of the reference signal, and digital means for dividing in multiple stages the frequency of the oscillator to that of the reference signal. The generator also comprises presettable logic means for detecting accumulation of count by the dividing means which corresponds to a selected phase angle difference between the variable and reference signals, and register means triggered by said logic means and adapted upon receipt of trigger therefrom to provide an output. The signal generator further comprises a subcarrier generator with an output of frequency higher than that of the reference signal, means for frequencymodulating said output with the output signal from the dividing means, means for amplitudemodulating said frequency-modulated output by the output signal from the register means, and means for applying the resulting composite signal to the VOR receiver to be tested. As described, the stable oscillator 10, Fig. 1A, is crystal controlled and has a frequency of 4À32 MHz, which is brought down to 60 Hz by division in five successive stages 12 to 16, having respective divisors of 4, 10, 10, 10, 18. The divider 16 has two outputs which are complementary in phase, and which are further divided by 2 in respective dividers 17, 18 to give outputs which have a frequency of 30 Hz and which are in phase quadrature. The output of the divider 17 constitutes the reference signal (f R ). Each stage of the dividers 13 to 16 corresponds, respectively, to 0À01, 0À1, 1 and 10 degrees of phase at 30 Hz. The required phase difference between the variable phase signal (f V ) and f R  in the range 0-180 degrees is preset in terms of those phase steps in respective comparators 21, 23, 24, 25 by means of a phase angle selector switch 22. Each comparator is connected by logic lines to a respective divider, and provides an input to an AND gate 27. For phase differences in the range 180-360 degrees the comparators are preset to accumulate phase increments totalling the difference between the selected phase angle and 180 degrees. A comparator 26 performs this function, receiving output from the divider 16, and control from the switch 22. A one-bit register 28 is toggled by pulses from the gate 27 to pass output from the divider 17, being a square wave signal f V  differing in phase from f R  by the amount selected on the switch 22. A logic circuit inverts the output of the register 28 for phase angle selections in the range 180- 360 degrees (Fig. 3, not shown). A switch 29 allows selection of a direct or inverted f V  output from the register 28 according to whether bearings to or from a VOR station are required, and said output is applied to a square wave to sine wave converter 31, Fig. 1B. The converter 31 provides one input to a summing amplifier 32, which acts as an amplitude modulator, either directly or through a voltage controlled phase shifter 33, as determined by the setting of a switch 65A. The f R  square wave output from the divider 17 is converted to sine wave form in a converter 34 and is applied as one input to a summing amplifier 37, which is comprised in a 9960 Hz subcarrier generator 35. The amplifier 37 also receives a control input from an integrator 43, and its output frequency modulates a 9960 Hz voltage controlled oscillator 36. A second input to the summing amplifier 32 is provided via a LP filter 38 by the VCO 36. Sine wave output from the filter 38 is converted to square wave form in a crossover detector 39 and is divided in frequency by a factor of 332 in a divider 41. Although the frequency of VCO 36 deviates 480 Hz, the average value thereof should remain at 9960 Hz. The output of divider 41 then should be a 30 Hz square wave in quadrature with the square wave from divider 17. The outputs of dividers 41 and 17 are compared in a quadrature phase detector 42 which produces a D.C. output having a magnitude and sense dependent upon the average frequency error of the VCO 36. Since the half-periods of a square wave cycle from divider 41 are unequal; i.e. the half-period resulting from dividing VCO output frequencies between 9960-10440 is shorter than the halfperiod resulting from dividing VCO output frequencies between 9960-9480, detector 42 is a symmetrical circuit which effectively averages the period of square wave from divider 41 in comparing it with the period of square wave from divider 17. The error signal from detector 42 is integrated and applied as a second control input to VCO 36 through summing amplifier 37. For test purposes, output from cross-over detector 39 is applied to a pulsewidth discriminator 44 which recovers the 30 Hz signal due to frequency modulation including any phase errors therein which have been introduced by converter 34, amplifier 37, VCO 36, filter 38, and detector 39. The demodulated reference signal from discriminator 44 is compared with variable signal from converter 31 in summing amplifier 45, filtered in filter 46 and detected in quadrature phase detector 47 which produces a D.C. output measurable on meter 48. Adjustment towards zero error may be made by a manual tuning control on the converter 31. The deviation ratio is measured in circuit 51, output from the divider 41 being supplied to a pulsewidth subtractor 52 which also receives f R  190 degrees from the divider 18. Because of the quadrature shift in phase resulting from frequency modulation, the square waves applied to subtractor 52 are in phase. However, the square wave, from divider 41 is unbalanced since divider 41 reaches a count of 166, corresponding to one-half cycle of output, more rapidly during the positive, or advancing frequency, half modulation cycle than during the negative, or retarding frequency, half modulation cycle. The difference in duration between the longer positive half cycle square wave from divider 18 and the shorter positive half cycle square wave from divider 41 is therefore a measure of the frequency deviation of VCO 36. This difference is evaluated by enabling an AND gate 53 for the time difference between the positive half cycle inputs to subtractor 52. 4À32 MHz clock pulses from oscillator 10 are thereupon passed to a binary counter 54. The count accumulated in counter 54 during the measurement period is read into a comparator which is preset to the binary equivalent of 2200 corresponding to a deviation ratio of 16. Comparator 55 controls a storage register 56 having indicating lamps 57, 58 which shows by the illumination of one or the other a high or low deviation ratio depending on whether the output of counter 54 is greater or less than 2200, or, by the illumination of both, a correct deviation ratio. A runout monitor 60 detects errors in phasing of the reference and variable signals of the composite signal originating internally of the generator for all values of selected bearing or the sum of such errors and any additional errors originating in an external r.f. signal generator 61 which the composite output of summing amplifier 32 modulates. A composite signal similar to the output of amplifier 32 is provided in summing amplifier 62 which combines 9960 Hz signals from filter 38 with 30 Hz variable signal from converter 31. The output of amplifier 62 differs from that of amplifier 32 only in that the latter includes a 1020 Hz signal from oscllator 63 which simulates the tone indentification used in VOR transmitters. R.F. generator 61 includes a demodulator from which composite signal is returned to the runout monitor through a buffer amplifier 64. A selector 65 connects monitor 60 either to amplifier 62 for measurement of internal error alone or to amplifier 64 for measurement of internal error plus the additional error introduced by the generator 61. The composite signal from the switch 65 is applied to a 9960 Hz limiter demodulator 66 which includes a pulsewidth discriminator similar to discriminator 44 preceded by a limiter amplifier. The output of demodulator 66, after filtering in filter 67, is a 30 Hz sine wave which, absent error, is precisely in quadrature with the square wave output of divider 17. Any departure between this phase relationship is detected in a quadrature phase detector 68 and presented as a direct voltage input proportional to the phase error to a summing amplifier 69. The 30 Hz a.m. component of composite signal at switch 65 is separated by a filter 71 and compared in a quadrature phase detector 72 with the 30 Hz f V  square wave from switch 29. Any difference between a quadrature phase relationship between these two signals produces a direct voltage proportional to the phase error which is also applied to summing amplifier 69. The output of amplifier 69 controls phase shifter 33 so as to introduce, when switch 65a is in the external position, a compensating phase shift in the f V  component, tending to drive the total error to zero. The output of amplifier 69 is also displayed on meter 48 which is calibrated to indicate the error of the system in hundredths of a degree. Detailed description of the 9960 Hz generator 35 (Fig. 2).-The phase detector 42 includes an input amplifier comprising complementary transistors 81, 82 wherein 30 Hz square wave f R  from divider 17 is converted to a bipolar square wave. The bipolar square wave is amplified in amplifiers 83, 84. Amplifier 84 is fed from the output of amplifier 83, thus providing at the outputs of these amplifiers two bipolar square waves, one of which is in phase with the reference signal, f R , and the other of which is 180